Abstract:
A method of transmitting data in a wireless communication system from a transmitter to a receiver, comprising the steps of modulating data at the transmitter using a first signal constellation pattern to obtain a first data symbol. The first data symbol is transmitteds to the receiver using a first diversity branch. Further, the data is modulated at the transmitter using a second signal constellation pattern to obtain a second data symbol. Then, the second data symbol is transmitted to the receiver over a second diversity path. Finally, the received first and second data symbol are diversity combined at the receiver. The invention further relates to a transmitter and a receiver embodies to carry out the method of invention.

Description:
This is a continuation of application Ser. No. 10/501,905 filed Oct. 20, 2004 now U.S. Pat. No. 7,164,727, the priority of which is claimed under 35 USC §120. Application Ser. No. 10/501,905 is a 371 of PCT/EP2002/011,695 filed Oct. 18, 2002. 

   FIELD OF THE INVENTION 
   The present invention relates generally to transmission techniques in wireless communication systems and in particular to a method, transceiver and receiver using transmit diversity schemes wherein the bit-to-symbol mapping is performed differently for different transmitted diversity branches. The invention is particularly applicable to systems with unreliable and time-varying channel conditions resulting in an improved performance avoiding transmission errors. 
   BACKGROUND OF THE RELATED ART 
   There exist several well known transmit diversity techniques wherein one or several redundancy versions relating to identical data are transmitted on several (at least two) diversity branches “by default” without explicitly requesting (by a feedback channel) further diversity branches (as done in an ARQ scheme by requesting retransmissions). For example the following schemes are considered as transmit diversity;
         Site Diversity: The transmitted signal originates from different sites, e.g. different base stations in a cellular environment.   Antenna Diversity: The transmitted signal originates from different antennas, e.g. different antennas of a multi-antenna base station.   Polarization Diversity: The transmitted signal is mapped onto different polarizations.   Frequency Diversity; The transmitted signal is mapped e.g. on different carrier frequencies or on different frequency hopping sequences.   Time Diversity: The transmitted signal is e.g. mapped on different interleaving sequences.   Multicode Diversity: The transmitted signal is mapped on different codes in e.g. a CDMA (Code Division Multiple Access) system.       

   There are known several diversity combining techniques. The following three techniques are the most common ones:
         a Selection Combining: Selecting the diversity branch with the highest SNR for decoding, ignoring the remaining ones.   Equal Gain Combining: Combining received diversity branches with ignoring the differences in received SNR.   Maximal Ratio Combining: Combining received diversity branches taking the received SNR of each diversity branch into account. The combining can be performed at bit-level (e.g. LLR) or at modulation symbol level.       

   Furthermore, a common technique for error detection/correction is based on Automatic Repeat request (ARQ) schemes together with Forward Error Correction (FEC), called hybrid ARQ (HARQ). If an error is detected within a packet by the Cyclic Redundancy Check (CRC), the receiver requests the transmitter to send additional information (retransmission) to improve the probability to correctly decode the erroneous packet. 
   In WO-02/067491 A1 a method for hybrid ARQ transmissions has been disclosed which averages the bit reliabilities over successively requested retransmissions by means of signal constellation rearrangement. 
   As shown therein, when employing higher order modulation formats (e.g. M-PSK, M-QAM with log 2 (M)&gt;2), where more than 2 bits are mapped onto one modulation symbol, the bits mapped onto a modulation symbol have different reliabilities depending on their content and depending on the chosen mapping. This leads for most FEC (e.g. Turbo Codes) schemes to a degraded decoder performance compared to an input of more equally distributed bit reliabilities. 
   In conventional communication systems the modulation dependent variations in bit reliabilities are not taken into account and, hence, usually the variations remain after combining the diversity branches at the receiver. 
   SUMMARY OF THE INVENTION 
   The object of the invention is to provide a method, transmitter and receiver which show an improved performance with regard to transmission errors. This object is solved by a method, transmitter and receiver as set forth in the independent claims. 
   The invention is based on the idea to improve the decoding performance at the receiver by applying different signal constellation mappings to the available distinguishable transmit diversity branches. The idea is applicable to modulation formats, where more than 2 bits are mapped onto one modulation symbol, since this implies a variation in reliabilities for the bits mapped onto the signal constellation (e.g. for regular BPSK and QPSK modulation all bits mapped onto a modulation symbol have the same reliability). The variations depend on the employed mapping and on the actually transmitted content of the bits. 
   Depending on the employed modulation format and the actual number of bits mapped onto a single modulation symbol, for a given arbitrary number (N&gt;1) of available diversity branches the quality of the averaging process is different Averaging in the sense of the present invention is understood as a process of reducing the differences in mean combined bit reliabilities among the different bits of a data symbol. Although it might be that only after using several diversity branches or paths a perfect averaging with no remaining differences is achieved, averaging means in the context of the document any process steps in the direction of reducing the mean combined bit reliability differences. Assuming on average an equal SNR for all available diversity branches, for 16-QAM 4 mappings (4 diversity branches) would be needed to perfectly average out the reliabilities for all bits mapped on any symbol. However, if e.g. only 2 branches are available a perfect averaging is not possible. Hence, the averaging should then be performed on a best effort basis as shown in the example below. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention will be more readily understood from the following detailed description of preferred embodiments with reference to the accompanying figures which show: 
       FIG. 1  an example for a 16-QAM signal constellation; 
       FIG. 2  an example for a different mapping of a 16-QAM signal constellation; 
       FIG. 3  two further examples of 16-QAM signal constellations; 
       FIG. 4  an exemplary embodiment of a communication system according to the present invention; 
       FIG. 5  details of a table for storing a plurality of signal constellation patterns; and 
       FIG. 6  shows the communication system according to the present invention with an interleaver/inverter section. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   The following detailed description is shown for a square 16-QAM with Gray mapping. However, without loss of generality the shown example is extendable to other M-QAM and M-PSK (with log 2 (M)&gt;2) formats. Moreover, the examples are shown for transmit diversity schemes transmitting an identical bit-sequence on both branches (single redundancy version scheme). Then again, an extension to a transmit diversity scheme transmitting only partly identical bits on the diversity branches can be accomplished. An example for a system using multiple redundancy versions is described in copending EP 01127244, filed on Nov. 16, 2001. Assuming a turbo encoder, the systematic bits can be averaged on a higher level as compared to the parity bits. 
   Assuming a transmit diversity scheme with two generated diversity branches, which are distinguishable at the receiver (e.g. by different spreading or scrambling codes in a CDMA system, or other techniques of creating orthogonal branches) and a transmission of the same redundancy version, usually the received diversity branches are combined at the receiver before applying the FEC decoder. A common combining technique is the maximal ratio combining, which can be achieved by adding the calculated log-likelihood-ratios LLRs from each individual received diversity branch. 
   The log-likelihood-ratio LLR as a soft-metric for the reliability of a demodulated bit b from a received modulation symbol r=x+jy is defined as follows: 
   
     
       
         
           
             
               
                 
                   LLR 
                   ⁡ 
                   
                     ( 
                     b 
                     ) 
                   
                 
                 = 
                 
                   ln 
                   ⁡ 
                   
                     [ 
                     
                       
                         Pr 
                         ⁢ 
                         
                           { 
                           
                             b 
                             = 
                             
                               1 
                               ⁢ 
                               
                                 | 
                               
                               ⁢ 
                               r 
                             
                           
                           } 
                         
                       
                       
                         Pr 
                         ⁢ 
                         
                           { 
                           
                             b 
                             = 
                             
                               0 
                               ⁢ 
                               
                                 | 
                               
                               ⁢ 
                               r 
                             
                           
                           } 
                         
                       
                     
                     ] 
                   
                 
               
             
             
               
                 ( 
                 1 
                 ) 
               
             
           
         
       
     
   
   As can be seen from  FIG. 1  (bars indicate rows/columns for which the respective bit equals 1), the mappings of the in-phase component bits and the quadrature component bits on the signal constellation are orthogonal (for M-PSK the LLR calculation cannot be simplified by separating into complex components, however the general procedure of bit-reliability averaging is similar). Therefore, it is sufficient to focus on the in-phase component bits i 1  and i 2 . The same conclusions apply then for q 1  and q 2 . 
   Assuming that Mapping  1  from  FIG. 1  is applied for the bit-to-symbol mapping for the 1 st  diversity branch, the log-likelihood-ratio LLR of the most significant bit (MSB) i 1  and the least significant bit (LSB) i 2  yields the following equations for a Gaussian channel: 
                   LLR   ⁡     (     i   1     )       =     ln   ⁡     [         ⅇ     -       K   ⁡     (     x   +     x   0       )       2         +     ⅇ     -       K   ⁡     (     x   +     x   1       )       2               ⅇ     -       K   ⁡     (     x   -     x   0       )       2         +     ⅇ     -       K   ⁡     (     x   -     x   1       )       2             ]               (   2   )                 LLR   ⁡     (     i   2     )       =     ln   ⁡     [         ⅇ     -       K   ⁡     (     x   -     x   1       )       2         +     ⅇ     -       K   ⁡     (     x   +     x   1       )       2               ⅇ     -       K   ⁡     (     x   -     x   0       )       2         +     ⅇ     -       K   ⁡     (     x   +     x   0       )       2             ]               (   3   )               
where x denotes the in-phase component of the normalized received modulation symbol r and K is a factor proportional to the signal-to-noise ratio. Under the assumption of a uniform signal constellation (x 1 =3x 0  regular 16-QAM) equations (2) and (3) can be fairly good approximated as shown in S. Le Goff, A. Glavieux, C. Berrou, “Turbo-Codes and High Spectral Efficiency Modulation,” IEEE SUPERCOMM/ICC &#39;94, Vol. 2 , pp. 645-649, 1994, and Ch. Wengerter, A. Golitschek Edler von Elbwart, E. Seidel, G. Velev, M. P. Schmitt, “Advanced Hybrid ARQ Technique Employing a Signal Constellation Rearrangement,” IEEE Proceedings of VTC 2002 Fall, Vancouver, Canada, September 2002 by
   LLR ( i   1 )=−4 Kx   0   x   (4)   LLR ( i   2 )=−4 Kx   0 (2 x   0   −|x| )  (5) 
   The mean LLR for i 1  and i 2  for a given transmitted modulation symbol yields the values given in Table 1 (substituting 4Kx 0   2  by Λ). Mean in this sense, refers to that the mean received value for a given transmitted constellation point, exactly matches this transmitted constellation point Individual samples of course experience noise according to the parameter K. However, for a Gaussian channel the mean value of the noise process is zero. In case of transmitted modulation symbols 0q 1  1q 2  and 1q 1  1q 2 , where q 1  and q 2  are arbitrary, the magnitude of the mean LLR (i i ) is higher than of the mean LLR (i 2 ). This means that the LLR for the MSB i 1  depends on me content of the LSB i 2 ; e.g. in  FIG. 1  i 1  has a higher mean reliability in case the logical value for i 2  equals 1 (leftmost and rightmost columns). Hence, assuming a uniform distribution of transmitted modulation symbols, on average 50% of the MSBs i 1  have about three times the magnitude in LLR of i 2 . 
   
     
       
             
           
             
             
             
             
             
           
         
             
               TABLE 1 
             
           
           
             
                 
             
             
               Mean LLRs for bits mapped on the in-phase component 
             
             
               of the signal constellation for Mapping 1 in FIG. 1 
             
             
               according to equations (4) and (5). 
             
           
        
         
             
                 
               Symbol 
               Mean 
                 
                 
             
             
                 
               (i 1 q 1 i 2 q 2 ) 
               value of x 
               Mean LLR (i 1 ) 
               Mean LLR (i 2 ) 
             
             
                 
                 
             
             
                 
               0q 1 0q 2   
               x 0   
                −4Kx 0   2  = −Λ 
               −4Kx 0   2  = −Λ 
             
             
                 
               0q 1 1q 2   
               x 1   
               −12Kx 0   2  = −3Λ 
                 4Kx 0   2  = Λ 
             
             
                 
               1q 1 0q 2   
               −x 0    
                4Kx 0   2  = Λ 
               −4Kx 0   2  = −Λ 
             
             
                 
               1q 1 1q 2   
               −x 1    
                 12Kx 0   2  = 3Λ 
                 4Kx 0   2  = Λ 
             
             
                 
                 
             
           
        
       
     
   
   If now adding a 2 nd  transmit diversy branch transmitting e.g. an identical bit sequence prior art schemes would employ an identical mapping to the 1 st  diversity branch. Here, it is proposed to employ a 2 nd  signal constellation mapping (Mapping  2 ) according to  FIG. 2  (of course, also one of the constellations depicted in  FIG. 3  are possible), which yields the mean LLRs given in Table 2. 
   
     
       
             
           
             
             
             
             
             
           
         
             
               TABLE 2 
             
           
           
             
                 
             
             
               Mean LLRs for bits mapped on the in-phase component 
             
             
               of the signal constellation for Mapping 2 in FIG. 2. 
             
           
        
         
             
                 
               Symbol 
               Mean 
               Mean 
               Mean 
             
             
                 
               (i 1 q 1 i 2 q 2 ) 
               value of x 
               LLR (i 1 ) 
               LLR (i 2 ) 
             
             
                 
                 
             
             
                 
               0q 1 0q 2   
               x 0   
               −Λ   
               −3Λ 
             
             
                 
               0q 1 1q 2   
               x 1   
               −Λ   
                 3Λ 
             
             
                 
               1q 1 0q 2   
               −x 0   
               Λ 
                −Λ 
             
             
                 
               1q 1 1q 2   
               −x 1   
               Λ 
                  Λ 
             
             
                 
                 
             
           
        
       
     
   
   Comparing now the soft-combined LLRs of the received diversity branches applying the constellation rearrangement (Mapping  1 + 2 ) and applying the identical mappings (Mapping  1 + 1 , prior art), it can be observed from table 3 that the combined mean LLR values with applying the constellation rearrangement have a more uniform distribution (Magnitudes: 4×4Λ and 4×2Λ instead of 2×6Λ and 6×2Λ). For most FEC decoders (e.g. Turbo Codes and Convolutional Codes) this leads to a better decoding performance. Investigations have revealed that in particular Turbo encoding/decoding systems exhibit a superior performance. It should be noted, that the chosen mappings are non exhaustive and more combinations of mappings fulfilling the same requirements can be found. 
   
     
       
             
           
             
             
             
           
             
             
             
             
           
             
             
             
             
             
             
           
         
             
               TABLE 3 
             
           
           
             
                 
             
             
               Mean LLRs (per branch) and combined mean LLRs for bits mapped 
             
             
               on the in-phase component of the signal constellation for the 
             
             
               diversity branches when employing Mapping 1 and 2 and when 
             
             
               employing 2 times Mapping 1. 
             
           
        
         
             
                 
               Constellation 
               Prior Art 
             
             
                 
               Rearrangement 
               No Rearrangement 
             
           
        
         
             
               Transmit 
                 
               (Mapping 1 + 2) 
               (Mapping 1 + 1) 
             
           
        
         
             
               Diversity 
               Symbol 
               Mean 
               Mean 
               Mean 
               Mean 
             
             
               Branch 
               (i 1 q 1 i 2 q 2 ) 
               LLR (i 1 ) 
               LLR (i 2 ) 
               LLR (i 1 ) 
               LLR (i 2 ) 
             
             
                 
             
             
               1 
               0q 1 0q 2   
               −Λ 
               −Λ 
               −Λ 
               −Λ 
             
             
                 
               0q 1 1q 2   
               −3Λ 
               Λ 
               −3Λ 
               Λ 
             
             
                 
               1q 1 0q 2   
               Λ 
               −Λ 
               Λ 
               −Λ 
             
             
                 
               1q 1 1q 2   
               3Λ 
               Λ 
               3Λ 
               Λ 
             
             
               2 
               0q 1 0q 2   
               −Λ 
               −3Λ 
               −Λ 
               −Λ 
             
             
                 
               0q 1 1q 2   
               −Λ 
               3Λ 
               −3Λ 
               Λ 
             
             
                 
               1q 1 0q 2   
               Λ 
               −Λ 
               Λ 
               −Λ 
             
             
                 
               1q 1 1q 2   
               Λ 
               Λ 
               3Λ 
               Λ 
             
             
               Combined 
               0q 1 0q 2   
               −2Λ 
               −4Λ 
               −2Λ 
               −2Λ 
             
             
               1 + 2 
               0q 1 1q 2   
               −4Λ 
               −4Λ 
               −6Λ 
               2Λ 
             
             
                 
               1q 1 0q 2   
               2Λ 
               −2Λ 
               2Λ 
               −2Λ 
             
             
                 
               1q 1 1q 2   
               4Λ 
               2Λ 
               6Λ 
               2Λ 
             
             
                 
             
           
        
       
     
   
   In the following an example with 4 diversity branches will be described. Here, the same principles apply as for 2 diversity branches. However, since 4 diversity branches are available and the averaging with 2 diversity branches is not perfect, additional mappings can be used to improve the averaging process, 
     FIG. 3  shows the additional mappings for diversity branches  3  and  4 , under the assumption that Mappings  1  and  2  are used for branches  1  and  2  (in  FIG. 1  and  FIG. 2 ). Then the averaging can be performed perfectly and all bits mapped on any symbol will have an equal mean bit reliability (assuming the same SNR for all transmissions). Table 4 compares the LLRs with and without applying the proposed Constellation Rearrangement. Having a closer look at the combined LLRs, it can be seen that with application of the Constellation Rearrangement the magnitude for all bit reliabilities results in 6Λ. 
   It should be noted again, that the chosen mappings are non exhaustive and more combinations of mappings fulfilling the same requirements can be found. 
   
     
       
             
           
             
             
             
           
             
             
             
             
           
             
             
             
             
             
             
           
         
             
               TABLE 4 
             
           
           
             
                 
             
             
               Mean LLRs (per branch) and combined mean LLRs for bits mapped 
             
             
               on the in-phase component of the signal constellation for the 
             
             
               diversity branches when employing Mappings 1 to 4 and when 
             
             
               employing 4 times Mapping 1. 
             
           
        
         
             
                 
               Constellation 
                 
             
             
                 
               Rearrangement 
               Prior Art 
             
             
                 
               (Mapping 
               No Rearrangement 
             
           
        
         
             
               Transmit 
                 
               1 + 2 + 3 + 4) 
               (Mapping 1 + 1 + 1 + 1) 
             
           
        
         
             
               Diversity 
               Symbol 
               Mean 
               Mean 
               Mean 
               Mean 
             
             
               Branch 
               (i 1 q 1 i 2 q 2 ) 
               LLR (i 1 ) 
               LLR (i 2 ) 
               LLR (i 1 ) 
               LLR (i 2 ) 
             
             
                 
             
             
               1 
               0q 1 0q 2   
               −Λ 
               −Λ 
               −Λ 
               −Λ 
             
             
                 
               0q 1 1q 2   
               −3Λ 
               Λ 
               −3Λ 
               Λ 
             
             
                 
               1q 1 0q 2   
               Λ 
               −Λ 
               Λ 
               −Λ 
             
             
                 
               1q 1 1q 2   
               3Λ 
               Λ 
               3Λ 
               Λ 
             
             
               2 
               0q 1 0q 2   
               −Λ 
               −3Λ 
               −Λ 
               −Λ 
             
             
                 
               0q 1 1q 2   
               −Λ 
               3Λ 
               −3Λ 
               Λ 
             
             
                 
               1q 1 0q 2   
               Λ 
               −Λ 
               Λ 
               −Λ 
             
             
                 
               1q 1 1q 2   
               Λ 
               Λ 
               3Λ 
               Λ 
             
             
               3 
               0q 1 0q 2   
               −Λ 
               −Λ 
               −Λ 
               −Λ 
             
             
                 
               0q 1 1q 2   
               −Λ 
               Λ 
               −3Λ 
               Λ 
             
             
                 
               1q 1 0q 2   
               Λ 
               −3Λ 
               Λ 
               −Λ 
             
             
                 
               1q 1 1q 2   
               Λ 
               3Λ 
               3Λ 
               Λ 
             
             
               4 
               0q 1 0q 2   
               −3Λ 
               −Λ 
               −Λ 
               −Λ 
             
             
                 
               0q 1 1q 2   
               −Λ 
               Λ 
               −3Λ 
               Λ 
             
             
                 
               1q 1 0q 2   
               3Λ 
               −Λ 
               Λ 
               −Λ 
             
             
                 
               1q 1 1q 2   
               Λ 
               Λ 
               3Λ 
               Λ 
             
             
               Combined 
               0q 1 0q 2   
               −6Λ 
               −6Λ 
               −4Λ 
               −4Λ 
             
             
               1 + 2 + 
               0q 1 1q 2   
               −6Λ 
               6Λ 
               −12Λ 
               4Λ 
             
             
               3 + 4 
               1q 1 0q 2   
               6Λ 
               −6Λ 
               4Λ 
               −4Λ 
             
             
                 
               1q 1 1q 2   
               6Λ 
               6Λ 
               12Λ 
               4Λ 
             
             
                 
             
           
        
       
     
   
   If the constellation rearrangement is performed by applying different mapping schemes, one would end up in employing a number of different mappings as given in  FIG. 1 ,  FIG. 2  and  FIG. 3 . If the identical mapper (e.g.  FIG. 1 ) should be kept for all transmit diversity branches, e.g. mapping  2  can be obtained from mapping  1  by the following operations:
         exchange positions of original bits i 1  and i 2      exchange positions of original bits q 1  and q 2      logical bit inversion of original bits i 1  and q 1          

   Alternatively, those bits that end in positions  1  and  2  can also be inverted (resulting in a different mapping with an identical bit-reliability characteristics). Accordingly, mapping  2  can be obtained from mapping  1 , using an interleaver/inverter section  14  (see  FIG. 6 ) which performs interleaving and/or inverting of the bits. 
   Therefore, the following table provides an example how to obtain mappings  1  to  4  (or mappings with equivalent bit reliabilities for i 1 , i 2 , q 1  and q 2 ), where the bits always refer to the first transmission, and a long dash above a character denotes logical bit inversion of that bit: 
   
     
       
             
           
             
             
           
         
             
               TABLE 5 
             
           
           
             
                 
             
             
               Alternative implementation of the Constellation Rearrangement 
             
             
               by interleaving (intra-symbol interleaving) and logical inversion 
             
             
               of bits mapped onto the modulation symbols. 
             
           
        
         
             
                 
               Interleaver and Inverter 
             
             
               Mapping No. 
               functionality 
             
             
                 
             
             
               1 
               i 1 q 1 i 2 q 2   
             
             
               2 
               ī 2   q   2 ī 1   q   1  or i 2 q 2 ī 1   q   1   
             
             
               3 
               ī 2   q   2 i 1 q 1  or i 2 q 2 i 1 q 1   
             
             
               4 
               i 1 q 1 ī 2   q   2  or ī 1   q   1 ī 2   q   2   
             
             
                 
             
           
        
       
     
   
   Generally at least 2 different mappings should be employed for N&gt;1 diversity branches, where the order and the selection of the mappings is irrelevant, as long as the bit-reliability averaging process, meaning the (reduction of differences in reliabilities) is maintained. 
   Preferred realizations in terms of number of employed mappings
         M-QAM
           Employing log 2 (M) different mappings   Employing log 2 (M)/2 different mappings   
           M-PSK
           Employing log 2 (M) different mappings   Employing log 2 (M)/2 different mappings   Employing 2 log 2 (M) different mappings   
               

   The applied signal constellation mappings for modulation at the transmitter and demodulation at the receiver need to match for each individual transmit diversity branch. This can be achieved by appropriate signaling of parameters indicating the proper mapping or combination of mappings to be applied for the diversity branches. Alternatively the definition of the mappings to be applied for transmit diversity branches may be system predefined. 
     FIG. 4  shows an exemplary embodiment of a communication system according to the present invention. More specifically, the communication system comprises a transmitter  10  and a receiver  20  which communicate through a communication channel consisting of a plurality of diversity branches  40 A,  40 B and  40 C. Although three diversity branches are illustrated in the figure, it becomes clear to a person skilled in the art that an arbitrary number of branches may be chosen. From a data source  11 , data packets are supplied to a FEC encoder  12 , preferably a FEC Turbo encoder, where redundancy bits are added to correct errors. The bits output from the FEC encoder are subsequently supplied to a mapping unit  13  acting as a modulator to output symbols formed according to the applied modulation scheme stored as a constellation pattern in a table  15 . Subsequently the data symbols are applied to a transmission unit  30  for transmission over the branches  40 A-C. The receiver  20  receives the data packets by the receiving unit  35 . The bits are then input into a demapping unit  21  which acts as a demodulator using the same signal constellation pattern stored in the table  15  which was used during the modulation of that symbol. 
   The demodulated data packets received over one diversity branch are stored in a temporary buffer  22  for subsequent combining in a combining unit  23  with the data packets received over at least one other diversity branch. 
   As illustrated in  FIG. 5 , table  15  stores a plurality of signal constellation patterns # 0  . . . #n which are selected for the individual transmissions over the individual diversity branches according to a predetermined scheme. The scheme, i.e. the sequence of signal constellation patterns used for modulating/demodulating are either pre-stored in the transmitter and the receiver or are signaled by transmitter to the receiver prior to usage.