Abstract:
A time domain reflectometry measuring instrument uses a microprocessor that provides added functionality and capabilities. The circuit electronics and probe are tested and calibrated at the factory. Installation and commissioning by the user is simple. The user installs the probe. The transmitter is attached to the probe. The user connects a standard shielded twisted pair to the electronics. Power is applied and the device immediately displays levels. A few-simple parameters may need to be entered such as output characteristics and the process material dielectric constant.

Description:
CROSS-REFERENCE 
     This is a Divisional of application Ser. No. 09/336,194, filed Jun. 18, 1999. 
     This application claims priority on Ser. No. 60/089,756 filed Jun. 18, 1998. 
    
    
     FIELD OF THE INVENTION 
     This invention relates to an apparatus and method utilizing time domain reflectometry for measuring the condition or characteristics of a material. 
     BACKGROUND OF THE INVENTION 
     Knowledge of level in industrial process tanks or vessels has long been required for safe and cost-effective operation of plants. Many technologies exist for making level measurements. These include buoyancy, capacitance, ultrasonic and microwave radar, to name a few. Recent advantages in micropower impulse radar (MIR), also known as ultra-wide band (UWB) radar, in conjunction with advances in equivalent time sampling (ETS), permit development of low power and low cost time domain reflectometry (TDR) devices. Existing TDR devices are expensive and often impractical for industrial level instrumentation. 
     In a TDR instrument, a very fast pulse with a rise time of 500 picoseconds, or less, is propagated down a transmission line that serves as a probe in a vessel. The pulse is reflected by a discontinuity caused by a transition between two media. For level measurement, that transition is typically where the air and the material to be measured meet. The amplitude of the reflected signal depends on the difference between the dielectrics of the two media. The dielectric of air is one, while the dielectric of water is about eighty. The larger the difference in dielectric, the larger the reflected signal. 
     McEwan, U.S. Pat. No. 5,345,471, and other related patents, describe a technique to transmit and receive very fast pulses with simple, low cost and low power electronics. More particularly, McEwan, U.S. Pat. No. 5,609,059, describes a level sensor utilizing this technology. However, the device described therein is intended to be used for simple commercial level applications, such as automobile engine fluid levels. It does not utilize the feature set, power consumption, and versatility required for use in the industrial process environment. Other known devices utilize this technology for a two-wire transmitter using just two wires for both receiving power from the user and sending level information to the user. However, these devices are analog devices limited in the ability to measure the level of extremely low dielectric materials, or materials that coat, clump or build up on the probe, over the wide temperature extremes of industrial process level environments. They also have limited level range capability. 
     The present invention is directed to overcoming one or more of the problems discussed above, in a novel and simple manner. 
     SUMMARY OF THE INVENTION 
     In accordance with the invention there is provided a TDR measurement instrument including improved diagnostic and measurement features. 
     In accordance with one aspect of the invention, there is disclosed a measurement instrument utilizing self diagnostics. 
     In accordance with another aspect of the invention, there is provided a TDR measurement instrument utilizing automated zoom scaling. 
     Broadly, there is disclosed herein a time domain reflectometry measurement instrument comprising a probe defining a transmission line. A pulse circuit is connected to the probe for generating pulses on the transmission line and receiving reflected pulses returned on the transmission line, the reflected pulses representing a characteristic of a material being measured. An equivalent time sampling circuit is connected to the pulse circuit for developing an equivalent time representation of the reflected pulses over an expanded time scale within the region of interest. An automatic scale circuit is operatively associated with the sampling circuit for automatically selecting location and size of the region of interest corresponding to characteristics of the probe. 
     More particularly, the instrument uses a microprocessor that enables added functionality and capabilities. The electronics and probe are tested and calibrated at the factory. Installation and commissioning by the user is simple. The user installs the probe. The transmitter is screwed on the top of the probe. The user connects a standard shielded twisted pair electrical cable to the electronics. Power is applied and the device immediately displays levels. A few simple parameters may need to be entered such as output characteristics and the process material dielectric constant. 
     Further features and advantages of the invention will be readily apparent from the specification and from the drawing. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a side elevation view of a measurement instrument in accordance with the invention; 
     FIG. 2 is a block diagram of a control circuit for the instrument of FIG. 1; 
     FIG. 3 is a timing diagram illustrating a return signal received by the circuit of FIG. 2; 
     FIG. 4 is a flow diagram illustrating operation of a program by the microprocessor of FIG. 2; 
     FIG. 5 is a block diagram of the ETS circuitry and MIR circuitry for the control circuit of FIG. 2; 
     FIG. 6 is a sectional view of an alternative probe that can be used with the instrument of FIG. 1 including a separation segment; and 
     FIG. 7 is a side view of the probe of FIG.  6 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring to FIG. 1, a process instrument  20  according to the invention is illustrated the process instrument  20  uses micropower impulse radar (MIR) in conjunction with equivalent time sampling (ETS) and ultra-wide band (UWB) transceivers for measuring level. Particularly, the instrument  20  uses guided wave radar for sensing level. While the embodiment described herein relates to an MIR level sensing apparatus, various aspects of the invention may be used with other types of process instruments for measuring various process parameters. 
     The process instrument  20  includes a control housing  22 , a probe  24 , and a connector  26  for connecting the probe  24  to the housing  22 . The probe  24  is typically mounted to a process vessel (not shown) using a threaded fitting  28 . The housing  22  is then secured to the probe  24  as by threading the connector  26  to the probe  24  and also to the housing  22 . These components may be as generally described in Mulrooney et al., U.S. patent application Ser. No. 09/094,142, filed Jun. 9, 1998, now U.S. Pat. No. 6,062,095, the specification of which is hereby incorporated by reference herein. As described therein, the probe  24  comprises a high frequency transmission line which, when placed in a fluid, can be used to measure level o fluid. Particularly, the probe  24  is controlled by electronics in the housing  22  for determining level in the vessel 
     Referring to FIG. 2, the electronics mounted in the housing  22  of FIG. 1 are illustrated in block diagram form as a control circuit  30  connected to the probe  24 . The control circuit  30  includes a microprocessor  32  connected to a suitable memory  34  (the combination forming a computer) and a display/pushbutton interface  36 . The display/push button interface  36  may be as generally described in the Mulrooney et al. application incorporated by reference herein, and is used for entering parameters with a keypad and displaying user information. The memory  34  comprises both non-volatile memory for storing programs and calibration parameters, as well as volatile memories used during level measurement, as described below. 
     The microprocessor  32  is also connected to digital to analog input/output circuitry  38 , which is in turn connected to two-wire 4-20 mA circuitry  40  for connecting to remote devices as represented by an input/output line  42 . Particularly, the two-wire circuitry  40  utilizes loop control and power circuitry which is well known and commonly used in process instrumentation. The power is provided on the line  42  from an external power supply. The circuitry  40  controls the current on the two-wire line  42  which represents level or other characteristics measured by the probe  24 . 
     The microprocessor  32  is also connected via logic and timing circuitry  44  to an ETS circuit  46 . The logic and timing circuitry  44  converts signals to appropriate levels and coordinates timing of such signal levels. The ETS circuit  46  is connected via an ME circuit  48  to the probe  24 . The ETS circuitry  46  and the MIR circuitry  48  are known and are generally in accordance with the teachings of McEwan, U.S. Pat. Nos. 5,345,471 and 5,609,059, the specifications of which are hereby incorporated by reference herein. The MIR circuitry  48  generates very fast pulses of 500 picoseconds or less rise time. The timing between pulses is tightly controlled. The reflected pulses are sampled at controlled intervals. The samples build a time multiplied “picture” of the reflected pulses. A typical picture is illustrated in FIG.  3 . All of the pulses shown in FIG. 3 will not be present at the same time. A clump pulse occurs only when the material measured forms a clump on the probe. When a level pulse is present typically there is no end of probe reflection. Since the pulses travel in the probe at the speed of light, this picture represents approximately ten nanoseconds in real time for a five-foot probe. The ETS circuitry  46  converts the time to about 71 milliseconds. As is apparent, the exact time would depend on various factors, such as, for example, probe length. The largest signals have an amplitude of the order of 20 millivolts before amplification to the desired amplitude by common audio amplifiers. For a low power device, a threshold scheme is employed to give timed interrupts to the microprocessor  32  for each of the signals, namely, fiducial, clump, level, and end of probe, see FIG.  3 . The microprocessor  32  converts these timed interrupts into distance. With the probe length entered through the display/pushbutton interface  36 , or some other interface, the microprocessor  32  can calculate the level by subtracting from the probe length the difference between the fiducial and level distances. 
     In accordance with the invention, the known ETS circuitry  46  is modified to include digital control by microprocessor  32  of selected circuit parameters. Here, three digital potentiometers labeled DP 1 , DP 2  and DP 3  receive digital commands, having one of 256 settings representing desired resistance, from the microprocessor  32 . The commanded resistance varies operation of the ETS circuitry  46  for controlling the signal sent via the logic and timing circuitry  44  to the microprocessor  32 . 
     Prior art level transmitters primarily utilize analog potentiometer adjustments for various probe lengths, or may have a single setting for the maximum probe length, relying on linearity of electronics to provide level measurement with a shorter probe. These approaches limit accuracy and resolution of a level measurement. They also create difficulties during field servicing of the level transmitters. If electronics must be replaced in the field, then all the calibrations and adjustments have to be made there as well, which requires more highly qualified field personnel, special test equipment, and more time to perform such procedures. In accordance with the invention, electronic zoom scaling allows the device to overcome these shortcomings. Electronic zoom scaling combines versatility, stability and universality of analog potentiometers with the capabilities of a microprocessor using the digital potentiometers DP 1  to DP 3 . The adjustments made by the respective digital potentiometers DP 1 , DP 2 , and DP 3  are range, window and gain. 
     Referring to FIG. 3, the curve illustrates the return signal received by the ETS circuitry  46 . The MIR circuitry  48  generates and transmits a measurement pulse at some known time to the left of the amplitude axis. The returned fiducial signal is a reference signal based on a marker built into the instrument  20 . Typically, the fiducial is positioned at some location along the probe, such as the top of the probe, higher than a highest level that can be sensed. The ETS circuitry  46  is configured to look for a return signal only in a region of interest (ROI). A window is defined as a time from when the pulse is transmitted to the beginning of the region of interest. The ETS circuitry  46  then looks for the received pulses only within the region of interest. The ETS circuitry  46  also utilizes an adjustable range. The range represents the length of the ROI beyond the end of the window to look for return signals. The first digital potentiometer DP 1  is used to adjust the range. The second digital potentiometer DP 2  is used to adjust the window. The range may be adjusted to include the fiducial signal and the end of probe signal. 
     The third potentiometer DP 3  is used for the gain adjustment. The gain adjustment controls the amplitude of the return signal as will appear below. The amount of reflected energy depends on the dielectric of the material. The gain adjustment controls the amplitude of the analog signal regardless of the media. 
     The window and range are influenced by the tolerances of the electronic components of a given circuit board. This relationship is determined during the factory calibration of the electronics. The relevant calibration parameters are stored in an onboard memory by factory personnel. If a board has to be replaced in the field, then a new one is installed and the pertinent setup parameters are entered. No additional calibration with test equipment, potentiometer adjustments, or level measurement is necessary. 
     The proper combination of range and window adjustment, in accordance with the invention, allows increased resolution and accuracy of level measurements by “zooming in” on the ROI. Such ROI may be an interface between air and liquid, or an interface between air and solids, or an interface between two liquids, or any other area along the probe. The microprocessor  32  and digital potentiometers DP 1 -DP 3  allow setting the location and size of the ROI and maintaining information about such selection with respect to the overall distance. 
     There is considerable interest in level measurement of materials with extremely low dielectric constants. Jet fuel and kerosene have a dielectric constant of the order of 1.7. Liquid nitrogen has a dielectric constant of the order of 1.45. The dielectric constant varies with temperature and with purity of the material. For these products, reflection coefficients are extremely small. By zooming in on a small ROI around the interface area, not only does the accuracy and resolution of the measurement increase, but it also allows for a lower sweep and therefore a larger number of pulses per inch. An increased number of pulses per inch increases the signal to noise ratio without increasing power consumption. Additionally, intentionally placed reference markers, or spacers, may be placed on the probe  24 . This provides absolute distance references which can be used to determine material level on the probe. 
     Referring to FIG. 4, a flow diagram illustrates a program implemented in the microprocessor  32  of FIG. 2 for level measurement. As discussed above, the instrument  20  is factory calibrated. Particularly, the factory adjusts a parameter referred to as “window offset” so that a known marker in the probe  24 , such as a fiducial, is located near the beginning of the ROI. The window offset value is stored in permanent memory for later recall. The factory then establishes a conversion factor which is a time multiplication factor representing distance per unit time, such as inches per second. The conversion factor is also stored in permanent memory. These calibrations reduce variations due to component tolerances. 
     The operation of the program in the microprocessor  32  begins at a start node  50 . Thereafter, the window offset value is read from memory  34  at a block  52 . The conversion factor is read from the memory  34  at a block  54 . Using the display/pushbutton interface  36 , the user enters the probe type, probe length and dielectric of the media at a block  56 . Thereafter, the program enters an automatic setup routine for controlling further operation. 
     At a block  58  the program uses probe type and probe length information to determine the location of markers intentionally designed into the probe to provide known reference points. As described herein, the instrument  20  uses two markers. However, the software can support any number of markers located at known points along the length of the probe  24 . The first is the fiducial at the top of the probe  24 . The second marker is the end of probe  24 . The end of probe  24  is an open circuit which generates a very high impedance. Calibration is done with no material covering the probe. Calibration of scaling parameters such as scale offset minimizes temperature and time drift of other parameters. Calibration is also used to set up the region of interest by determining the window and range. 
     The software automatically adjusts range to cover the distance between the two or more markers. This is done by either formulas or lookup table using empirically determined data at a block  60 . The ETS circuitry  46  is updated by writing the range value to the range digital potentiometer DP 1 . At a block  62 , the program automatically adjusts a window adder so that the desired marker is at the beginning of the ROI. A block  64  calculates a window value. The window value is equal to the sum of the window offset, read at the block  52 , and the window adder calculated at the block  62 . The ETS circuitry  46  is updated by writing the window value to the window digital potentiometer DP 2 . The window adder is determined either by formula or look-up table. 
     At a block  66 , the program adjusts the conversion factor and scale offset based on the value of range calculated at the block  60  and the probe information entered by the user. This is also done either by formula or lookup table. Finally, a block  68  calculates the gain based on the probe type and dielectric information entered by the user. This is done by formula or lookup table. The ETS circuitry  46  is updated by writing the gain value to the gain digital potentiometer DP 3 . 
     Thereafter, a decision block  70  determines if it is necessary to return to the setup routine. Normally, the setup routine, comprising the blocks  52 - 68 , is implemented only if a change is made by a user entering parameters using the display/pushbutton interface  36  at the block  56 . Otherwise, the program remains in the level measurement routine. 
     The level measurement routine begins at a block  72 , where a pulse is generated. As is apparent, the generation of the pulse actually consists of a repetitive pulse sequence using the ETS and MIR technologies. A decision block  74  determines if there has been a fiducial pulse, as shown in FIG.  3 . If the fiducial pulse is received, decision block  75  determines if a return level pulse is received within the ROI after the fiducial. If so, then a block  76  determines level. This is done by measuring the time between the fiducial marker at the beginning of the ROI and the reflection from the media. The time is multiplied by the conversion factor and added to the scale offset to obtain the distance from the top of the probe to an interface of dissimilar materials. The level is the probe length minus the distance plus an offset, which is the distance from the end of the probe to a predetermined reference point. The routine then returns to the decision block  70 . The level measurement routine is repeated unless the user has changed a parameter, in which event the setup routine is repeated. 
     If a return pulse is not received, as determined at the decision block  75 , then a self diagnostic routine is entered. Particularly, an expected time of flight is known between the fiducial and the end of probe. If no signal is received within that time, then the instrument  20  is not operating properly. A decision block  77  determines if an end of probe signal is detected within the expected time limit. If so, then the control returns to the decision block  70  and the level measurement routine is repeated unless the user has changed a parameter, in which event the setup routine is repeated. 
     If the fiducial signal or, in the absence of a level pulse, the end of probe signal is missing, an error condition is indicated at a block  78 . An error message is given at display interface  36  and the current on two-wire line  42  is at an error level, e.g., less than 4 mA or more than 20 mA. The control returns to block  70 , as described above. 
     Thus, for normal operation, the instrument  20  continues to measure level as long as the fiducial signal and a return level pulse or an end of probe pulse are received during an expected time period. However, when a setup parameter is changed, then the control returns to the setup routine for adjusting the electronic zoom scaling parameters. 
     As discussed above, the ETS circuitry  46  is modified to include the digital potentiometers DP 1 -DP 3 . Referring to FIG. 5, a block diagram illustrates the modified ETS circuitry  46  and the MIR circuitry  48  in block diagram form. 
     A pulse repetition frequency (PRF) oscillator  83  is connected to a transmit pulse shaper  84  which is in turn connected to the probe  24 . The oscillator  83  is also connected to a pulse input of a voltage controlled pulse delay generator  85 . Pulse outputs of the delay generator  85  and the oscillator  83  are also connected to a time delay to voltage converter  86 . The output of the delay generator  85  is also connected to a receive pulse shaper  87  that is connected to a sampler  88 . The sampler  88  is connected to the probe. An output of the sampler  88  is connected to a signal amplifier  89  which includes the gain digital potentiometer DP 3 . The signal amplifier  89  is in turn connected to a signal processing circuit  90  which transfers the return pulse signal to the microprocessor  32  of FIG.  2 . 
     A sawtooth ramp generator  91  develops a sawtooth ramp as an input to a summing amplifier  92 . The summing amplifier  92  is connected to the range potentiometer DP 1  and the window potentiometer DP 2 . Range potentiometer DP 1  sets gain, while Window potentiometer DP 2  sets a DC offset applied to the ramp signal from the ramp generator  91 . The output of the summing amplifier  92  is supplied as an input to a difference amplifier  93 . The other input to the difference amplifier  93  is an output from the time delay to voltage converter  86 . These input signals represent respectively a desired delay and an actual delay. The output of the difference amplifier is an error signal which is supplied to a control voltage input of the pulse delay generator  85 . 
     The PRF oscillator  83  generates a pulse train at the frequency of the transmitted pulses. For each input pulse, the pulse delay generator  85  generates an output pulse with a delay that is controlled by its control voltage input. The delayed pulse passes through the receive pulse shaper  87  and is provided to the sampler  88 . The sampler  88  samples the reflected pulses from the probe  24  at a time determined by the delayed and shaped receive pulse. The resulting low frequency sampled signal is amplified and passed on to further signal processing circuits  89  and  90 . The amplifier gain is controlled by the computer controlled gain digital potentiometer DP 3 . 
     The PRF oscillator signal and the delayed pulse also are inputs to the time delay to voltage converter  86 . The output of this converter  86  is a voltage proportional to the actual time delay between the transmit pulse from the PRF oscillator  83  and the receive pulse from the pulse delay generator  85  The voltage representing the measured delay is subtracted from a voltage representing the desired delay by the difference amplifier  93 . The difference amplifier  93  output is an error signal that controls the time delay in the voltage controlled pulse delay generator  85 . This feedback loop measures the actual pulse delay, compares it to the desired pulse delay, and forces the voltage controlled pulse delay generator  85  to adjust the actual pulse delay until it matches the desired delay. In this way, most nonlinearities and drifts in the voltage controlled pulse generator  85  have little or no effect on the actual pulse delay. 
     The voltage representing the desired delay is produced by the sawtooth ramp generator  91  and the summing amplifier  92 . The summing amplifier amplifies the sawtooth ramp signal and adds a DC offset to the resulting signal. The DC offset voltage and the gain of the summing amplifier are controlled by the respective computer controlled digital potentiometers DP 2  and DP 1 . 
     Automated control of the window (DC offset) and range (sawtooth ramp gain) using potentiometers DP 1  and DP 2  allows the microprocessor  32  to control the starting voltage and ending voltage of the sawtooth ramp representing the desired time delay, and therefore to control through the feedback loop the starting and ending time delay values actually applied to the sampler  88 . Controlling these time delays, in turn, controls the distance down the probe  24  that the scan begins and ends, allowing the microprocessor  32  to scan any small or large portion of the probe as may be needed to easily identify a level reflection from the probe  24 . All spurious signals, as reflections occurring from locations outside the scanned interval, are not seen by the signal processing circuits. 
     Microprocessor control of the gain potentiometer DP 3  allows for adaptation to differing signal levels caused by differing dielectric constants of the reflecting medium and for differing signal amplitudes resulting from differing probe configurations. 
     The probe  24  may be a coaxial probe including a signal conductor surrounded by a ground sleeve. Such a coaxial probe and the cables exhibit an impedance of approximately 50 ohms. Another type of probe is a twin rod probe. This type of probe has a mismatch, typically beyond the fiducial. In accordance with the invention, a separation segment is used to provide impedance matching in the form of a gradual impedance change from the fiducial to the twin lead line. This enables the instrument to measure small dielectrics without interference from characteristics of the probe itself. 
     Referring to FIG. 6, a twin rod probe  100  according to the invention is illustrated. The probe  100  includes mounting means, as a flange  102  or thread. A tubular fitting  104  extends outwardly from the flange  102 . The fitting  104  has an outer threaded end  106  and is shown with a cap  108 . In use, the cap  108  is removed so the fitting can threadably receive the instrument housing  22 . A bushing  110  is welded to an inside of the flange  102  and has a through opening  111  coaxial with a flange opening  109  and the fitting  104 . A transmission line conductor  112  is centered in the fitting  104  and bushing opening  111  with a spacer  114 . A coaxial connector  115  is provided at an outer end of the conductor  112 . A separation sleeve  116  is connected inwardly of the bushing  110 , coaxial with and receiving the conductor  112 . An additional Teflon spacer  118  is provided in the separation sleeve  116 . 
     The probe  100  also includes a signal line  120  and a ground line  122  connected in parallel with spacers  124 . The signal line  120  is secured to the conductor  112 . The ground line  122  is fastened to the bushing  110 . In accordance with the invention, the probe  100  includes a fiducial section  126  representing the area of the conductor  112  outwardly of the signal line  120  and ground line  122 , a separation segment  128 , and a level measuring segment  130 . The level measuring segment  130  corresponds to location of of the signal line  120 . 
     A coaxial cable (not shown) which is connected between the MIR circuitry  48  and probe  100  has an impedance of 50 ohms. The center conductor of the coaxial cable is connected through connector  115  with conductor  112 . Fiducial section  126  has an input impedance of 50 ohms, matching the coaxial cable. Conductor  112  has a narrowed portion  134  within the fiducial section  126 . This narrowed portion  134  has a greater spacing from flange  102  and bushing  110  than the remainder of conductor  112  and thus has a greater impedance, of the order of 75 ohms. The 25 ohm mismatch causes the fiducial return signal of FIG.  3 . The output impedance of conductor  112 , at the end of bushing  110 , is also 50 ohms. 
     A coaxial probe (not shown) having an impedance of 50 ohms would match the impedance of fiducial section  126  and be connected directly to the end thereof. However, coaxial probes cannot be used in some conditions, as with the measurement of materials of high viscosity, which would tend to clog the space between the inner and outer conductors. 
     The twin rod probe shown in FIGS. 6 and 7 has a higher impedance than that of fiducial section  126 , typically of the order of 150 ohms. Separation segment  128  provides an impedance match between fiducial section  126  and level measuring segment  130 . Separation sleeve  116  surrounding conductor  112  beyond bushing  110  has an angled cut  132  which provides a gradual impedance change from 50 ohms to 150 ohms. This reduces reflections to the MIR circuitry  48  from the transition between conductor  112  and signal line  120 . Teflon spacer  118  provides a stable mechanical connection between conductor  112  and signal line  120  and prevents liquids from entering fiducial section  126 . 
     The control circuit  30  has the capability of digital communication through the two-wire circuitry  40  and line  42  with remote devices and the outside world. Such communication preferably uses the HART protocol. 
     Thus, in accordance with the invention, there is described a TDR measurement instrument which uses zoom scaling for improved measurement, self diagnostics, and improved probe design for certain applications.