Abstract:
A method and system for wireless communication is provided and may include mitigating blocker signals in transmitted RF signals in a wireless device including a transmitter front end. The mitigation may include up-converting a baseband signal, mixing the up-converted baseband signal with a feedback signal, and amplifying the mixed up-converted baseband signal and the feedback signal to generate an output signal. The feedback signal may be generated by down-converting the output signal, low-pass filtering the down-converted output signal, and up-converting the filtered down-converted signal for the summing. The blocker signals may correspond to receive frequencies for the wireless device and may be converted to DC via the down-converting. The wireless devices may operate in accordance with at least a CDMA standard. The mixed up-converted baseband signal and the feedback signal may be amplified utilizing a source follower amplifier, where the gain and/or linearity may be configured by varying coupling capacitors.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS/INCORPORATION BY REFERENCE 
     This application is a continuation of U.S. application Ser. No. 11/833,048 filed Aug. 2, 2007. 
    
    
     FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT 
     [Not Applicable] 
     MICROFICHE/COPYRIGHT REFERENCE 
     [Not Applicable] 
     FIELD OF THE INVENTION 
     Certain embodiments of the invention relate to wireless communication. More specifically, certain embodiments of the invention relate to a highly linear and very low-noise down-conversion mixer for extracting weak signals in the presence of very strong unwanted signals. 
     BACKGROUND OF THE INVENTION 
     In some conventional systems, a transmitter may transmit radio frequency (RF) signals via direct up-conversion or indirect up-conversion. During direct up-conversion, the RF signals may be generated by directly up-converting the baseband signals to RF signals. During indirect up-conversion, the RF signals are generated by up-converting baseband signals to intermediate frequency (IF) signals, and then further up-converting the IF signals to RF signals. The baseband signals may be up-converted to RF signals by using local oscillator (LO) signals that may be at a desired RF carrier frequency. The baseband signal, or the IF signal, may be mixed with the LO signal to generate the RF signal. However, amplifying the RF signal in preparation for transmission may allow a relatively strong RF transmit signal to leak to RF receiver circuitry, where portions of the RF signal may provide blocking for the received RF signals. For example, the RF transmit signal may comprise frequencies that may interfere with received RF signals. 
     Further limitations and disadvantages of conventional and traditional approaches will become apparent to one of skill in the art, through comparison of such systems with some aspects of the present invention as set forth in the remainder of the present application with reference to the drawings. 
     BRIEF SUMMARY OF THE INVENTION 
     A system and/or method is provided for a highly linear and very low-noise down-conversion mixer for extracting weak signals in the presence of very strong unwanted signals, substantially as shown in and/or described in connection with at least one of the figures, as set forth more completely in the claims. 
     Various advantages, aspects and novel features of the present invention, as well as details of an illustrated embodiment thereof, will be more fully understood from the following description and drawings. 
    
    
     
       BRIEF DESCRIPTION OF SEVERAL VIEWS OF THE DRAWINGS 
         FIG. 1  is a block diagram of an exemplary wireless system, which may be utilized in connection with an embodiment of the invention. 
         FIG. 2A  is a diagram illustrating an exemplary frequency spectrum for RF signals in a transmit front-end of a wireless system. 
         FIG. 2B  is a diagram illustrating an exemplary frequency spectrum for RF signals in a transmit front-end of a wireless system after being down-converted to baseband. 
         FIG. 3A  is a block diagram of an exemplary noise canceling circuit, which may be utilized in connection with an embodiment of the invention. 
         FIG. 3B  is a block diagram of an exemplary circuit for extracting a desired signal, which may be utilized in connection with an embodiment of the invention. 
         FIG. 4A  is a diagram of an exemplary highly linear, low-noise circuit that may be used to extract a desired, weak signal in the presence of strong signals, in accordance with an embodiment of the invention. 
         FIG. 4B  is a simplified diagram that shows an equivalent circuit for a portion of the exemplary highly linear, low-noise circuit shown in  FIG. 4A . 
         FIG. 4C  is a simplified diagram that shows an equivalent circuit for a portion of the exemplary highly linear, low-noise circuit shown in  FIG. 4A . 
         FIG. 4D  is an exemplary diagram that shows further filtering provided for the exemplary highly linear, low-noise circuit shown in  FIG. 4A , in accordance with an embodiment of the invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Certain embodiments of the invention may be found in a highly linear and very low-noise down-conversion mixer for extracting weak signals in the presence of very strong unwanted signals. Exemplary aspects of the invention may comprise a source follower circuit, in a transmitter front end of, for example, a mobile terminal, which may receive RF signals prior to the RF signals being amplified by a power amplifier for transmission. The RF signals may comprise in-phase and quadrature components. The source follower circuit may be enabled to generate output RF voltage signals, and communicate the output RF voltage signals to a switching circuit via a coupling capacitor. The switching circuit may be enabled to down-convert the communicated output RF voltage signals to generate differential baseband signals. The capacitance of the coupling capacitor may be changed to trade gain and/or linearity to the noise of the system. 
     The mobile terminal may operate in accordance with, for example, a CDMA standard. Accordingly, the mobile terminal may transmit at a frequency that is 190 MHz below a corresponding receive frequency. The switching circuit may, for example, down-convert to DC the receive frequency for the mobile terminal. Each of the differential baseband signals may be low-pass filtered to attenuate higher frequency components including portions of the strong transmitter signal. 
       FIG. 1  is a block diagram of an exemplary wireless system, which may be utilized in connection with an embodiment of the invention. Referring to  FIG. 1 , a mobile terminal  100  may comprise a transmitting antenna  101 , a transmitter/receiver switch  101   a , a transmitter front end  102 , a receiver front end  103 , a baseband processor  104 , a processor  106 , and a system memory  108 . The transmitter/receiver switch  101   a  may comprise suitable circuitry that enables the antenna  101  to be used for both receiving and transmitting signals. The transmitter front end  102  may comprise suitable logic, circuitry, and/or code that may be adapted to upconvert baseband signals directly to RF signals and to transmit the RF signals via a transmitting antenna  101 . The transmitter front end  102  may also be adapted to upconvert a baseband signal to an IF signal, and/or upconvert the IF signal to an RF signal and then transmit the RF signal via the transmitting antenna  101 . The transmitter front end  102  may be adapted to execute other functions, for example, filtering the baseband signal, amplifying the baseband signal, filtering RF signals, and/or amplifying RF signals. 
     The receiver front end  103  may comprise suitable logic, circuitry, and/or code that may be adapted to downconvert a RF signal directly to a baseband signal for further processing. The receiver front end  103  may also be adapted to downconvert a RF signal to an IF signal, and/or downconvert the IF signal to a baseband signal for further processing. The receiver front end  103  may be adapted to execute other functions, for example, filtering the baseband signal, and/or amplifying the baseband signal. 
     The baseband processor  104  may comprise suitable logic, circuitry, and/or code that may be adapted to process baseband signals, for example, convert a digital signal to an analog signal, and/or vice-versa. The processor  106  may be any suitable processor or controller such as a CPU or DSP, or any type of integrated circuit processor. The processor  106  may comprise suitable logic, circuitry, and/or code that may be adapted to control the operations of the transmitter front end  102  and/or the baseband processor  104 . For example, the processor  106  may be utilized to update and/or modify programmable parameters and/or values in a plurality of components, devices, and/or processing elements in the transmitter front end  102  and/or the baseband processor  104 . Control and/or data information, which may include the programmable parameters, may be transferred from at least one controller and/or processor, which may be part of the mobile terminal  100 , to the processor  106 . Similarly, the processor  106  may be adapted to transfer control and/or data information, which may include the programmable parameters, to at least one controller and/or processor, which may be part of the mobile terminal  100 . 
     The processor  106  may utilize the received control and/or data information, which may comprise the programmable parameters, to determine an operating mode of the transmitter front end  102 . For example, the processor  106  may be utilized to select a specific frequency for a local oscillator, or a specific gain for a variable gain amplifier. Moreover, the specific frequency selected and/or parameters needed to calculate the specific frequency, and/or the specific gain value and/or the parameters needed to calculate the specific gain, may be stored in the system memory  108  via the processor  106 . The information stored in system memory  108  may be transferred to the transmitter front end  102  from the system memory  108  via the processor  106 . The system memory  108  may comprise suitable logic, circuitry, and/or code that may be adapted to store a plurality of control and/or data information, including parameters needed to calculate frequencies and/or gain, and/or the frequency value and/or gain value. In one embodiment of the invention, the processor  106  may be enabled to dynamically control operation of the transmitter front-end  102 . 
       FIG. 2A  is a diagram illustrating an exemplary frequency spectrum for RF signals in a transmit front-end of a wireless system. Referring to  FIG. 2A , there is shown a curve  200  of an exemplary frequency distribution of transmit RF signals generated by, for example, the transmitter front end  102 . The frequency  202  may be a nominal center frequency for transmission by, for example, a CDMA mobile terminal, such as, for example, the mobile terminal  100 . While the transmit RF signals may have been filtered to reduce signal strength outside the transmission bandwidth, there may still be frequencies where unwanted signals, or noise, exist. For example, the frequency  204  may be a center frequency for a receive channel for the mobile terminal  100 . To conform, for example, to the CDMA standards, the frequency  204  may be 190 MHz above the frequency  202 . 
     While the signal level at frequency  204  may be attenuated with respect to signal level at frequency  202 , the noise at frequency  204  may leak to the receiver front end  103  due to finite isolation of the duplexer, such as, for example, the transmitter/receiver switch  101   a . Accordingly, the signals received by the antenna  101  and communicated to the receiver front end  103  may be contaminated with the leaked noise from the transmitter front end  102 . Accordingly, the receiver front end  103  may encounter more errors in processing the received signals than if there were no leaked signals from the transmitter front end  102 . 
       FIG. 2B  is a diagram illustrating an exemplary frequency spectrum for RF signals in a transmit front-end of a wireless system after being down-converted to baseband. Referring to  FIG. 2B , there is shown a curve  210  of an exemplary frequency distribution of transmit RF signals generated by, for example, the transmitter front end  102 , after the transmit RF signals shown with respect to  FIG. 2A  has been down-converted using a local oscillator signal whose frequency may be the frequency  204 . Accordingly, the noise at frequency  204  may be centered about DC  212 , while the transmit center frequency may be offset by 190 MHz. 
     The noise centered about DC  212  may be attenuated by filtering without too much adverse effect on the signals of interest for transmission about frequency  214 , or 190 MHz. The filtered signals, which may comprise the relatively unaffected signals about the frequency  214 , and attenuated signals about DC  212 , may then be up-converted using a local oscillator signal whose frequency may be the frequency  204 . 
       FIG. 3A  is a block diagram of an exemplary noise canceling circuit, which may be utilized in connection with an embodiment of the invention. Referring to  FIG. 3A , there is shown a mixer  300 , signal combiner  302 , a pre-power amplifier (PA) amplifier  304 , a circuit block  310 , and a power amplifier  320 . The mixer  300  may comprise suitable circuitry that may enable up-conversion of baseband signals to RF signals. The signal combiner  302  may comprise suitable circuitry that may enable combining of output TX 1  from the mixer  300  with output TX 3  from the circuit block  310 . If the outputs TX 1  and TX 3  are currents, the signal combiner  302  may, for example, wire-OR the output TX 1  and TX 3 . 
     The pre-PA amplifier  304  may comprise suitable logic and/or circuitry that may enable amplification of the output signal TX 2  from the signal combiner  302 . The circuit block  310  may comprise suitable circuitry that may enable extraction of noise signal, for example, the noise signal about the frequency  204 . The power amplifier  320  may comprise suitable logic and/or circuitry that may enable amplification of RF signals for transmission via an antenna, such, as for example, the antenna  101 . 
     In operation, an input signal TX-IN, which may be a baseband signal, may be up-converted to RF by the mixer  300 , where the mixer  300  may use a local oscillator signal that may be a transmit RF center frequency. The output of the mixer  300  may be referred to as TX 1 . The RF signal TX 1  may be combined with the output signal TX 3  from the circuit block  310  to generate the RF signal TX 2 . The RF signal TX 2  may be amplified by the pre-PA amplifier  304  to generate the RF signal TX-OUT. The RF signal TX-OUT may be, for example, further amplified and/or filtered by the transmitter front end  102  before being transmitted via the antenna  101 . For example, the RF signal TX-OUT may be amplified by the power amplifier  320 , the output of the power amplifier  320  may be communicated to the antenna  101  for transmission. 
     The RF signal TX-OUT may also be processed by the circuit block  310  to, for example, extract the RF noise about the RF frequency  204 , which may be referred to as TX 3 . The RF noise TX 3  may be combined with the RF signal TX 1  to attenuate the noise about the RF frequency  204 . 
     While single signals, such as, for example, TX-IN, TX 1 , TX 2 , TX 3 , TX-OUT, may have been shown for illustrative purposes, the circuitry in  FIG. 3A  may comprise, for example, in-phase (I) and quadrature (Q) components of the various signals, and/or differential I and Q signals. 
       FIG. 3B  is a block diagram of an exemplary circuit for extracting a desired signal, which may be utilized in connection with an embodiment of the invention. Referring to  FIG. 3B , there is shown the circuit block  310  that comprises mixers  312  and  316 , and a low-pass filter  314 . The mixer  312  may mix the RF signal TX-OUT with a local oscillator signal LO-RX, where the frequency of the local oscillator signal LO-RX may be the frequency  204 . This may result in generating a baseband signal TX 4  comprising noise about DC and signals to be transmitted about the frequency  214 . 
     The baseband signal TX 4  may be filtered by the low-pass filter  314  to generate the noise signal TX 5 . The noise signal TX 5  may be those signals that, when up-converted to transmission frequencies, may be about the receive frequency  204 . The noise signal TX 5  may then be mixed by the mixer  316  with the local oscillator signal LO-RX, where the frequency of the local oscillator signal LO-RX may be the frequency  204 . The output of the mixer  316  may be the RF signal TX 3 . While single signals, such as, for example, TX 3 , TX 4 , TX 5 , and TX-OUT, are shown for illustrative purposes, the circuitry in  FIG. 3B  may comprise, for example, in-phase (I) and quadrature (Q) components of the various signals, and/or differential I and Q signals. 
       FIGS. 4A-4D  may describe circuitry that may be used to extract weak signals in the presence of strong unwanted signals, where the circuitry performs a highly linear and very low-noise down-conversion and appropriate filtering of the down-converted signal. 
       FIG. 4A  is a diagram of an exemplary highly linear, low-noise circuit that may be used to extract a desired, weak signal in the presence of strong signals, in accordance with an embodiment of the invention. Referring to  FIG. 4 , there is shown transistors  400 ,  402 ,  404 ,  406 ,  408 ,  410 , and  412 , resistors  414 ,  418 , and  424 , and capacitors  416 ,  420 , and  422 . The transistors  402  and  404  may comprise, for example, a switching circuit  403 . The resistor  418  and the capacitor  420  may comprise, for example, a filter  406   a . Similarly, the resistor  424  and the capacitor  422  may comprise, for example, a filter  408   a.    
     The transistors  400 ,  402 ,  404 ,  406 ,  408 ,  410 , and  412  may be, for example, NMOS transistors. The transistor  400  may be configured as a source follower where the output voltage may follow, for example, the input voltage. Accordingly, a source follower circuit  400   a  may comprise the transistor  400  and the resistor  414 . The gate terminal of the transistor  400  may receive an input signal, which may be, for example, the RF signal TX-OUT described with respect to  FIG. 3B . The drain terminal of the transistor  400  may be directly connected to a power source VDD, and the source terminal of the transistor  400  may be coupled to ground via the resistor  414 . The source terminal of the transistor  400  may also provide an output voltage signal, which may follow the input voltage signal. Accordingly, the transistor  400  may act as a buffer that may not introduce non-linearities due to, for example, saturation. 
     The transistor  400  may be coupled to a first terminal of the capacitor  416 , and a second terminal of the capacitor  416  may be coupled to source terminals of the transistors  402  and  404 . Accordingly, the AC coupled signal at the sources of the transistors  402  and  404  may not have a DC current. Since there is no DC current, there may not be flicker noise generated by the transistors  402  and  404 . The gates of the transistors  402  and  404  may have as inputs local oscillator signals LOp and LOn, which may be differential local oscillator signals. The differential local oscillator signals LOp and LOn may be used to down-convert, for example, the frequency of the input signal Vin, which may be AC coupled to the transistors  402  and  404 , to baseband signals at the drain terminals of the transistors  402  and  404 . 
     The baseband signals VD 1  and VD 2  may be output by the drain terminals of the transistors  402  and  404 , respectively. These signals may be referred to as, for example, the baseband signal TX 4 , and may be communicated to the source terminals of the transistors  406  and  408 , respectively, and to the drain terminals of the transistors  410  and  412 , respectively. The baseband signals output by the drain terminals of the transistors  402  and  404  may be, for example, the baseband signal TX 4 , as described with respect to  FIG. 3B . The transistors  410  and  412  may be configured as current sources for the transistors  406  and  408 , respectively. The source terminals of the transistors  410  and  412  may be coupled to ground, and the gate terminals of the transistors  410  and  412  may receive a common input signal, for example, Vb 1 . The voltage of the common input signal Vb 1  may be design and/or implementation dependent. 
     The transistors  406  and  408  may provide buffering and/or gain for the baseband signals generated by the transistors  402  and  404 . The gate terminals of the transistors  406  and  408  may be provided a common input signal, for example, Vb 2 . The voltage of the common input signal Vb 2  may be design and/or implementation dependent. The drain terminals of the transistors  406  and  408  may output signals OUT+ and OUT−, which may have been collectively referred to as, for example, the noise signal TX 5  with respect to  FIG. 3B . 
     The transistors  406 ,  408 ,  410 , and  412  may be fabricated sufficiently large in size so that they may not generate enough flicker noise to noticeably affect the current flowing through the transistors. The amount of flicker noise allowed and the size of the transistors  406 ,  408 ,  410 , and  412  may be design dependent. 
     The filters  406   a  and  406   b  may attenuate higher frequencies of the output signals OUT+ and OUT− at the drain terminals of the transistors  406  and  408 . Accordingly, the attenuated higher frequencies of the output signals OUT+ and OUT− may comprise the frequencies of the RF signal TX-OUT that may be about the frequency  214 . This may allow the signal strength of the output signals OUT+ and OUT−, or the noise signal TX 5 , to be more predominant at the lower frequencies about DC. The noise signal TX 5  may be up-converted to generate the output signal TX 3 , as described with respect to  FIG. 3B , and the output signal TX 3  may be combined with the RF signal TX-OUT. Accordingly, the noise frequencies about the frequency  204  may be attenuated while minimizing attenuation of the signals about the transmit frequency  202 . 
       FIG. 4B  is a simplified diagram that shows an equivalent circuit for a portion of the exemplary highly linear, low-noise circuit shown in  FIG. 4A . Referring to  FIG. 4B , there is shown a circuit that comprises resistors  430  and  434 , and a capacitor  432 . The resistor  430  may represent, for example, an equivalent resistance seen at the source terminal of the transistor  400 . The capacitor  432  may be, for example, the capacitor  416  that AC couples the output of the transistor  400  to the transistors  402  and  404 . The resistor  434  may represent, for example, an equivalent resistance of the transistor  406  at the source terminal of the transistor  406 . The baseband signal VD 1  may be the output signal at the drain terminal of the transistor  410 , which may be communicated to the source terminal of the transistor  406 . 
     Accordingly, the voltage of the baseband signal VD 1  may be described by the following equation:
 
 VD 1=( Vin )*( Rm 2)/( Rm 1 +Rm 2− jXc ).
 
RM 1  may be the resistance of the resistor  430 , RM 2  may be the resistance of the resistor  434 , and −jXc may be the impedance of the capacitor  432 . Accordingly, since the voltage of the baseband signal VD 1  may be smaller in value than the voltage of the input signal Vin, the output of the transistor  406  OUT+ may not have any issues related to linearity due to, for example, saturation of an amplifying transistor.
 
     A similar description may also apply to the portion of the circuitry involving the transistors  400 ,  404 , and  408 , and the capacitor  416 . Accordingly, the output voltage VD 2  may be described by the following equation:
 
 VD 2=( Vin )*( Rm 3)/( Rm 1 +Rm 3 −jXc ),
 
where Rm 3  may be the equivalent resistance of the transistor  408  at the source terminal of the transistor  408 . Since the transistors  406  and  408  may be fabricated to be similar, the resistance Rm 3  may be similar to the resistance Rm 2 . Accordingly, the amplitude of the baseband signal VD 1  may be equal to the amplitude of baseband signal VD 2 . The ratio of VD 1 /Vin may be design and/or implementation dependent.
 
       FIG. 4C  is a simplified diagram that shows an equivalent circuit for a portion of the exemplary highly linear, low-noise circuit shown in  FIG. 4A . Referring to  FIG. 4C , there is shown a current source  440 , a resistor  442 , and a capacitor  444 . The current source  440  may comprise, for example, a current ID 1 , which may be the current through the transistor  406 , and the output signal OUT+. The resistor  442  and the capacitor  444  may comprise, for example, the filter  406   a . Accordingly, the current ID 1  may be filtered by the filter  406   a , thereby attenuating higher frequency components from the output signal OUT+. The current though the transistor  408  may be similarly filtered by the filter  408   a , where the output signal OUT− may also comprise attenuation of higher frequency components. The attenuation at specific frequencies may be dependent on the values of the resistor  442  and the capacitor  444 . These values may be design and/or implementation dependent. 
       FIG. 4D  is an exemplary diagram that shows further filtering provided for the exemplary highly linear, low-noise circuit shown in  FIG. 4A , in accordance with an embodiment of the invention. Referring to  FIG. 4D , there is shown the capacitors  450  and  452  that may be coupled to the drain terminals of the transistors  402  and  404 , respectively. The capacitors  402  and  404 , in conjunction with the equivalent resistances of the transistors  406  and  408 , may form a filter that may attenuate high frequency portions of the baseband signals VD 1  and VD 2 . 
     In accordance with an embodiment of the invention, aspects of an exemplary system may comprise the source follower circuit  400   a  in the transmitter front end  102  of the mobile terminal  100 . The source follower circuit  400   a  may receive RF signals prior to the RF signals being amplified for transmission by the power amplifier  320 . The RF signals may comprise in-phase and quadrature components. A coupling capacitor  416  may communicate output RF voltage signals generated by the source follower circuit  400   a  to the switching circuit  403 , where the switching circuit may comprise the transistors  402  and  404 . The switching circuit  403  may generate differential output signals. The capacitance of the coupling capacitor  416  may be changed to control gain and/or linearity of the differential output signals. 
     The capacitors  450  and  452  may be coupled to the outputs of the transistors  402  and  404 , respectively, to each of the differential outputs of the switching circuit  403 . The capacitors  450  and  452  may act in conjunction, for example, with the input resistances at the source terminals of the transistors  406  and  408  to attenuate high frequency signals. The frequencies that may be filtered may be design and/or implementation dependent. 
     The mobile terminal  100  may operate, for example, in accordance with at least a CDMA standard. Accordingly, the switching circuit  403  may down-convert the output RF voltage signals, communicated from the transistor  400  via the transistor  416 , to baseband signals. The switching circuit  403  may use, for example, local oscillator signals that may allow down-conversion of the frequency  204  to DC. The frequency  204  may be, for example, the receive frequency for a CDMA mobile terminal. 
     Accordingly, the present invention may be realized in hardware, software, or a combination of hardware and software. The present invention may be realized in a centralized fashion in at least one computer system, or in a distributed fashion where different elements are spread across several interconnected computer systems. Any kind of computer system or other apparatus adapted for carrying out the methods described herein is suited. A typical combination of hardware and software may be a general-purpose computer system with a computer program that, when being loaded and executed, controls the computer system such that it carries out the methods described herein. 
     The present invention may also be embedded in a computer program product, which comprises all the features enabling the implementation of the methods described herein, and which when loaded in a computer system is able to carry out these methods. Computer program in the present context means any expression, in any language, code or notation, of a set of instructions intended to cause a system having an information processing capability to perform a particular function either directly or after either or both of the following: a) conversion to another language, code or notation; b) reproduction in a different material form. 
     While the present invention has been described with reference to certain embodiments, it will be understood by those skilled in the art that various changes may be made and equivalents may be substituted without departing from the scope of the present invention. In addition, many modifications may be made to adapt a particular situation or material to the teachings of the present invention without departing from its scope. Therefore, it is intended that the present invention not be limited to the particular embodiment disclosed, but that the present invention will comprise all embodiments falling within the scope of the appended claims.