Abstract:
A technique, which substantially reduces the number of power-stage and control circuit components in an isolated DC/DC converter with parallel current-doubler rectifier stages, includes on the primary side transformers with serially connected primary windings each having a corresponding secondary winding coupled to one of the voltage-doubler stages. In one embodiment, the primary and secondary windings and filter inductors of the current-doubler rectifier stages are provided on an integrated magnetic core. The filter inductors in each current-doubler rectifier stage can be provided as coupled inductors. In one embodiment, an X-shaped magnetic core is provided to achieve coupled or uncoupled filter inductors.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates to isolated dc/dc converters. In particular, this invention relates to low output voltage, high output current, isolated dc/dc converters that has multiple rectifier stages connected in parallel. 
     2. Discussion of the Related Art 
     In a high-power application, by connecting several substantially identical converter power stages in a parallel configuration to share the total power processed, one can often achieve a desired output power using smaller, lower-rated magnetic and semiconductor components. With several power stages connected in parallel, the power losses and thermal stresses on the magnetic and semiconductor components are distributed among the parallel power stages, thus improving conversion efficiency and eliminating “hot spots”. In addition, because lower-power, faster semiconductor switches can be used to implement the parallel power stages, the parallel power stages may be operated at a higher switching frequency than that of a corresponding single high-power stage. Consequently, the parallel configuration reduces the required sizes of the magnetic components and increases conversion power density. In addition, because the parallel power stages can be operated at a higher switching frequency, this approach can be used to optimize the transient response of a power supply. 
     FIG. 1 shows converter  100  with two forward-converter power stages  101  and  102  connected in parallel. Generally, a power supply with parallel power stages requires more power stage and control circuit components. However, if the parallel converters share the same output filter, the number of power stage components can be reduced, such as illustrated by converter  200  of FIG.  2 . Similarly, if transformer secondary windings are provided directly in parallel, required power stage components can also be reduced, such as illustrated by converter  300  of FIG.  3 . Converters  200  and  300  of FIGS. 2 and 3 are discussed in “Analysis, Design, and Evaluation of Forward Converter with Distributed Magnetics—Interleaving and Transformer Paralleling,” (“Zhang”) by M. T. Zhang, M. M. Jovanovic and F. C. Lee, published in  IEEE Applied Power Electronics Conf . ( APEC )  Proc ., pp. 315-321, 1995. 
     Regardless of the approach used in connecting power stages in parallel, ensuring that an acceptable load current (hence, power) is shared among the parallel modules is the main design challenge of such an approach. In fact, without an acceptable current-sharing mechanism, the load current can be unevenly distributed among the parallel modules. As a result, the modules that carry higher currents are electrically and thermally stressed more than the other modules, thus reducing the reliability of the power supply. Moreover, when the current of a parallel module exceeds its current limit, such as may occur when the converter current is unevenly distributed, the entire power supply may need to be shut off. Therefore, many current-sharing techniques of different complexities and performance are developed to ensure a relatively even current distribution among parallel modules. A discussion of some of these techniques is found in “A Classification and Evaluation of Paralleling Methods for Power Supply Modules,” by S. Luo, Z. Ye, R. L. Lin, and F. C. Lee, published in  IEEE Power Electronics Specialists&#39; Conf. Rec ., pp. 901-908, 1999. For example, relatively even current sharing in converters  100  and  200  in FIGS. 1 and 2 can be achieved by equalizing the peak values of primary currents in the modules. Furthermore, the performance of converter  100  and  200  of FIGS. 1 and 2 can be further improved by interleaving (i.e., operating the primary switches in each converter with 180° phase shift). Generally, as discussed by Zhang above, interleaving provides some input current and output current ripple cancellation, thus reducing the size of the input and output filters. 
     Referring to FIG. 3, steady-state current sharing among parallel transformers  301  and  302  of converter  300  is determined by the winding resistances of transformers  301  and  302 . Because winding resistance is usually comparable with the layout resistance, the current sharing performance of parallel transformers is sensitive to circuit layout. Sensitivity to layout resistance can be reduced by including a rectifier in the secondary side of each transformer, such as shown in converter  400  of FIG.  4 . In converter  400 , current sharing is determined by the on-resistances of rectifiers  401  and  402 , as a rectifier&#39;s resistance is usually larger than that of a printed circuit board (PCB) trace resistance. However, because the on-resistance of silicon rectifiers has a negative temperature coefficient (i.e., the rectifier&#39;s resistance decreases as the temperature of the rectifier increases), a current runaway condition may exist. In a runaway condition, all the secondary current flows through one of the rectifiers and the associated transformer secondary windings. The runaway condition in converter  400  can be avoided if low on-resistance MOSFETs (which have positive on-resistance temperature coefficients) are used instead of the diode rectifiers, as it is routinely done in low-voltage high-current applications. 
     In a low output voltage (e.g., below 3.3 V), high output current (e.g., above 50 A) application that requires transformer isolation, secondary-side conduction loss dominates total loss and limits conversion efficiency. Therefore, to increase conversion efficiency, rectification and transformer winding losses must be reduced. Rectification loss can be reduced, for example, by replacing Schottky rectifiers with MOSFET synchronous rectifiers. Reduction of transformer winding loss can be achieved by reducing winding resistance and the root-mean-square (rms) current in the winding, respectively, by properly selecting the winding geometry and transformer structure, and by employing a current-doubler topology. These techniques are discussed for example in “Design and Performance Evaluation of Low-Voltage/High-Current Dc/Dc On-Board Modules,” (“Panov”) by Y. Panov, M. M. Jovanovic, published in  IEEE Applied Power Electronics Conf . ( APEC )  Proc ., pp. 545-552, 1999, and in “The Performance of the Current Doubler Rectifier with Synchronous Rectification,” by L. Balogh, published in  High Frequency Power Conversion Conf. Proc ., pp. 216-225, 1995. 
     FIG. 5 shows an example of a 1.45-volt, 70-ampere dc/dc converter  500  that employs a current-doubler topology implemented with synchronous rectifiers. (Converter  500  is discussed in the Panov reference mentioned above). In converter  500 , synchronous rectifier  501  and  502  are each implemented by connecting three low on-resistance MOSFETs in parallel. The technique used in converter  500 , however, cannot be extended to higher current levels by simply adding more synchronous rectifier MOSFETs, because the incremental reduction in conduction losses is less than the incremental increase of switching losses due to charging and discharging of MOSFETs&#39; relatively large intrinsic terminal capacitances. If the switching frequency were not reduced, conversion efficiency would be reduced. However, reduction of switching frequency requires an undesirable increase in the sizes of magnetic components. In addition, the packaging of a large number of paralleled synchronous rectifiers is also difficult. 
     The output current of converter  500  of FIG. 5 can be increased without efficiency degradation by connecting in parallel two or more power stages, as illustrated in converter  600  of FIG.  6 . However, converter  600  requires significantly more power-stage and control circuit components to achieve even current (hence, power) sharing among the parallel modules. The additional components increase both the size and the cost of the converter. 
     SUMMARY OF THE INVENTION 
     According to the present invention, a parallel technique, which substantially reduces the number of power-stage and control-circuit components in an isolated dc/dc converter with a current-doubler rectifier and provides automatic current sharing is described. Using a common primary side inverter, and by providing in parallel only the secondary-side current-doubler rectifiers that are driven through separate isolation transformers, component count reduction is achieved. Current sharing among the parallel rectifier stages is achieved by connecting the primary windings of the transformers in series, thus forcing the same current through the transformers&#39; secondary windings and the rectifiers. Additional component count reduction is achieved using integrated magnetic components. The technique of the present invention can be extended to an arbitrary number of rectifier stages, as well as to any rectifier topology. 
     The present invention is better understood upon consideration of the following detailed description and the accompanying drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 shows prior art converter  100  having two forward converter stages  101  and  102  connected in parallel. 
     FIG. 2 shows prior art converter  200  having two forward converter stages  201  and  202  connected in parallel and sharing a common output filter. 
     FIG. 3 shows prior art forward converter  300  having transformers  301  and  302  connected in parallel upstream to rectifier  303 . 
     FIG. 4 shows prior art forward converter  400  having transformers  403  and  404  connected in parallel downstream from rectifiers  401  and  402 . 
     FIG. 5 shows prior art half-bridge converter  500  having a current-doubler output stage implemented by synchronous rectifiers  501  and  502 . 
     FIG. 6 shows prior art half-bridge converters  601  and  602 , each having a current-doubler rectifier, connected in parallel. 
     FIG. 7 shows, schematically, dc/dc converter  700 , having an arbitrary number N of parallel rectifier stages, according to one embodiment of the invention. 
     FIG. 8 shows key waveforms of converter  700  of FIG. 7, including (a) output voltage V inv  of inverter  701 ; (b) secondary voltage V si , representative of a secondary voltage at one of transformers  709 - 1  to  709 -N; (c) voltage V L1i , representative of a voltage across one of filter inductors  703 - 1  to  703 -N; (d) voltage V L2i , representative of a voltage across one of filter inductors  704 - 1  to  704 -N; (e) primary current i PRIM ; (f) currents i L1i  and i L2i , representative of the respective currents in one of filter inductors  703 - 1  to  703 -N and in one of filter inductors  704 - 1  to  704 -N; (g) current i D1i , representative of a current in one of rectifiers  705 - 1  to  705 -N; (h) current i D2i , representative of a current in one of rectifiers  706 - 1  to  706 -N. 
     FIG. 9 shows converter  900  using magnetic coupling of output filters, in accordance with a second embodiment of the present invention. 
     FIG. 10 shows an implementation of converter  900  of FIG. 9, using integrated magnetic components that has no magnetic coupling between filter inductors of the same rectifier stage. 
     FIG. 11 shows another implementation of converter  900  of FIG. 9, using integrated magnetic components that has magnetic coupling between filter inductors of the same rectifier stage. 
     FIG. 12 shows converter  1200  with rectifiers  1201  and  1202 , having integrated magnetic components on a single magnetic core (separation of core halves  1203 - 1  and  1203 - 2  is exaggerated for clarity). 
     FIG. 13 shows a model of the magnetic reluctance circuit of converter  1200  of FIG.  12 . 
     FIG. 14 shows converter  1300 , which is an alternative implementation of converter  1200  of FIG. 12, using opposite winding orientations to reduce the magnetic flux through center post  1203 - 3 . Note that the orientation of windings (dot positions) on legs  2  and  3  are opposite to the orientation of the corresponding windings in FIG.  12 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     In the detailed description below, to facilitate illustration and correspondence between figures, like elements are provided like reference numerals. 
     FIG. 7 shows, schematically, dc/dc converter  700  that has an arbitrary number N of parallel rectifier stages  707 - 1  to  707 -N, according to one embodiment of the invention. Dc/dc converter  700  uses inverter  701  to convert the dc input signal into a bipolar high-frequency square-wave signal that is applied across the series connection of primary windings  702 - 1  to  702 -N of transformers  709 - 1  to  709 -N. Inverter  701  can be implemented by virtually any converter topology, such as a forward converter, a two-switch forward converter, a half-bridge converter, or a full-bridge converter. As shown in FIG. 7, converter  700  has secondary windings  708 - 1  to  708 -N of transformers  709 - 1  to  709 -N each coupled to a respective one of current-doubler rectifiers  707 - 1  to  707 -N. Current-doubler rectifiers  707 - 1  to  707 -N are connected in parallel at the output terminals of converter  700 . Of course, rectifiers  705 - 1  to  705 -N and  706 - 1  to  706 -N can be implemented by synchronous rectifiers, such as those discussed above with respect to FIG.  5 . 
     Because primary windings  702 - 1  to  702 -N of transformers  709 - 1  to  709 -N are connected in series, a common current i PRIM  flows in all primary windings  702 - 1  to  702 -N (assuming that the primary windings of transformers  709 - 1  to  709 -N have identical magnetizing inductances). Consequently, if each pair of corresponding primary and secondary windings has the same turns ratio, secondary currents i SEC  in each of secondary windings  708 - 1  to  708 -N are also the same, which ensures a perfect current (hence, power) sharing among rectifier stages  707 - 1  to  707 -N. However, if the magnetizing inductances are different, secondary currents i SEC  will also be different. Because the variation of magnetizing inductance can be easily kept within a narrow range, variations in magnetizing inductances do not significantly affect current sharing. 
     FIG. 8 shows representative key waveforms of converter  700  of FIG.  7 . It should be noted that in FIG. 8 the symmetrical bipolar high-frequency voltage waveform at the output of the inverter implies that a symmetrical inverter topology (bridge-type topology) is assumed in the analysis that follows. 
     Ideally, when all components of rectifier stages  707 - 1  to  707 -N are identical, the waveforms of signals in rectifier stages  707 - 1  to  707 -N are identical. Thus, under ideal conditions, perfect current sharing is achieved, so that each rectifier stage carries 1/N of total load current i LOAD . Under ideal conditions, primary voltage V pi  across each of primary windings  702 - 1  to  702 -N is  1 /N input voltage V, or: 
     
       
         V P1 =V P2 = . . . =V Pn =V/N 
       
     
     Initially, as shown in FIG. 8 between time t 0  to t 1 , voltage V INV  of inverter  701  (magnitude V) is applied equally across each of primary windings  702 - 1  to  702 -N, thus inducing positive voltage V si =n*V/N across each of secondary windings  708 - 1  to  708 -N, where n is the turns ratio across each corresponding pair of primary and secondary windings. (FIGS.  8 ( a ),  8 ( b )) Consequently, rectifiers  705 - 1  to  705 -N are each in an “off” state (FIG.  8 ( g )), carrying no appreciable current. At the same time, a positive voltage V L1i  develops across each of inductors  703 - 1  to  703 -N (FIG.  8 ( c )), thus increasing inductor current i L1i  (FIG.  8 ( f )), which flows in the loop consisting of corresponding secondary windings  708 - 1  to  708 -N, rectifier  706 - 1  to  706 -N and filter capacitor  710 - 1  to  710 -N. Because rectifiers  706 - 1  to  706 -N conduct (FIG.  8 ( h )), voltage V L2i  across inductors  704 - 1  to  704 -N is negative and equals in magnitude to output voltage V o  (FIG.  8 ( d )). As a result, inductor current i L2i  in each of inductor  704 - 1  to  704 -N is linearly decreasing (FIG.  8 ( f )). 
     Between time t 1  and t 2  (i.e., time interval [t 1 , t 2 ]), voltage V INV  of inverter  701  is zero (FIG.  8 ( a )), inductor current i L1i  in each of inductors  703 - 1  to  703 -N, which was flowing during time interval [t 0 , t 1 ] through corresponding secondary windings  708 - 1  to  708 -N, continues to flow through rectifiers  705 - 1  to  705 -N (FIGS.  8 ( f ) and  8 ( g )). During time interval [t 1 , t 2 ], voltage V L1i  or V L2i  (FIGS.  8 ( c ) and  8 ( d )) across each inductor—i.e., any of inductors  703 - 1  to  703 -N and  704 - 1  to  704 -N—is negative and equal to output voltage V o . Consequently, current i L1i  or i L2i  in each inductor is decreasing linearly at the same rate (FIG.  8 ( f )). 
     During time intervals [t 2 , t 3 ] and [t 3  and t 4 ], the output voltage V INV  of inverter  701  is negative and zero, respectively. During these time intervals, the operations of converter  700  are identical to those of time intervals [t 0 , t 1 ] and time intervals [t 1 , t 2 ], except that the roles of inductors  703 - 1  to  703 -N and rectifiers  705 - 1  to  705 -N are exchanged with those of inductors  704 - 1  to  704 -N and rectifiers  706 - 1  to  706 -N. 
     In rectifier stages  707 - 1  to  707 -N, because voltage V L1i  across each of inductors  703 - 1  to  703 -N is the same, inductors  703 - 1  to  703 -N can be coupled, such as illustrated by coupled inductor  901  of converter  900  in FIG.  9 . (Similarly, because voltage V L2i  across each of inductors  704 - 1  to  704 -N is the same, inductors  704 - 1  to  704 -N can be coupled, such as also illustrated by coupled inductor  902  of converter  900  in FIG. 9) Using coupled inductors  901  and  902 , the number of magnetic cores required to implement output filtering is reduced to two. Further reduction of the magnetic core count can be achieved by integrating coupled inductors  901  and  902  of FIG. 9 onto a single magnetic core, such as illustrated in FIG. 10 for converter  1000  with two converter stages. Of course, the same concept can be extended to any number of rectifier stages. In the integrated magnetic implementation of converter  1000  in FIG. 10, outer legs of EE core  1003  are gapped where the windings of coupled inductors  901  and  902  are placed. As shown in FIG. 10, the center leg of EE core  1003  has no gap and, therefore, has a much lower reluctance than the gapped outer legs. As a result, any flux generated in either of the outer legs is closed through the center leg (i.e., no coupling exists between opposite windings, so that there is no interaction between inductors  703 - 1  and  703 - 2  on one outer leg of EE core  1003  with inductors  704 - 1  and  704 - 2  on the other outer leg of EE core  1003 ). 
     Alternatively, the magnetic integration of output filters can be also implemented by allowing a certain degree of coupling between filter inductors  703 - 1  and  703 - 2  wound on one leg of an EE core, and filter inductors  704 - 1  and  704 - 2  wound on the other leg of the EE core, as illustrated by EE core  1101  of converter  1100 , shown in FIG.  11 . In FIG. 11, the coupling between inductors  703 - 1 ,  703 - 2  and inductors  704 - 1  and  704 - 2  wound on two outside legs of EE core  1101  is achieved by gapping the middle leg of EE core  1101 . Due to an increased reluctance of the gapped middle leg of EE core  1101 , relative to EE core  1003  of FIG. 10, some flux that is generated in one outer leg of EE core  1101  is forced to flow in the other outer leg of EE core  1101 , thus coupling all windings of inductors  703 - 1 ,  703 - 2 .  704 - 1  and  704 - 2 . When a proper amount of coupling is provided, the ripple in filter inductors  703 - 1 ,  703 - 2 ,  704 - 1  and  704 - 2  of converter  1100  is less than the corresponding filter inductors in converter  1000  of FIG. 10, thus improving converter performance. 
     Converter  900  of FIG. 9 can also be implemented using a single magnetic core, such as illustrated by converter  1200  of FIG.  12 . In converter  1200 , 4-legged X-type magnetic core  1203  is used. Note that, for illustrative purpose, core halves  1203 - 1  and  1203 - 2  are shown in FIG. 12 with an exaggerated separation. Actual separation between core halves  1203 - 1  and  1203 - 2  is typically a few millimeters, or less. In FIG. 12, core halves  1203 - 1  and  1203 - 2  implement coupled filter inductors  703 - 1 ,  703 - 2 ,  704 - 1 , and  704 - 2  in the legs labeled “ 1 ” and “ 2 ”. Transformer windings  702 - 1 ,  702 - 2 ,  708 - 1  and  708 - 2  are implemented on the legs labeled “ 3 ” and “ 4 ”. To ensure correct operation of converter  1200 , magnetic core  1203  is properly gapped, so that the fluxes created by the transformer windings are provided in the desired magnetic paths. To illustrate the gapping requirements, FIG. 13 shows reluctance circuit  1300  that models the magnetic structure of core  1203  of FIG.  12 . 
     Generally, in an implementation such as converter  1200  of FIG. 12, a magnetic coupling between the transformers and the filter inductors is not desired. Because filter inductors are intended to store energy, legs  1  and  2  of EE core  1203  are gapped to create relatively large reluctances R 1  and R 2 , which are represented in FIG. 13 by respective reluctances  1303  and  1306 . In FIG. 13, inductors  703 - 1  and  703 - 2  in leg  1  of EE core  1203  are represented by voltage sources  1301  and  1302 , respectively. Similarly, inductors  704 - 1  and  704 - 2  in leg  2  of core  1203  are represented in FIG. 13 by voltage sources  1305  and  1304 . Because the transformers in converter  1200  are not intended to store energy, legs  3  and  4  need not be gapped. Reluctances in legs  3  and  4  are represented in FIG. 13 by reluctances  1312  and  1309 , respectively. However, without a gap, reluctances R 3  and R 4  are relatively small (i.e., reluctance R 3  and R 4  would each be comparable to reluctance R c  of non-gapped center post  1203 - 3 , which is represented in FIG. 13 by reluctance  1313 ). Primary windings  702 - 1  and  702 - 2  are represented in FIG. 13 by voltage sources  1307  and  1310 , respectively. Similarly, secondary windings  708 - 1  and  708 - 2  are represented in FIG. 13 by voltage sources  1308  and  1311 . As a result of the relative reluctances of the transformers to those of the inductors, a part of fluxes Φ 1  and Φ 2  produced by inductor currents in legs  1  and  2  of core  1203  would flow through legs  3  and  4 , in addition to the part of fluxes Φ 1  and Φ 2  flowing through center post  1203 - 3 . The amount of this flux coupling between the transformer legs and the inductor legs depends on the ratio of reluctance R 3  or reluctance R 4  to center-post reluctance R c . To minimize this coupling, reluctances R 3  and R 4  should be made much larger than reluctance R c  by not having a gap in center post  1203 - 3 , and by introducing small gaps in legs  3  and  4 . The gaps in legs  3  and  4  are generally much smaller than the gaps in legs  1  and  2 . In addition, when the air gaps are designed to achieve R c &lt;&lt;R 3 =R 4 &lt;&lt;R 1 =R 2 , flux linkage between legs  3  and  4  is also minimized (i.e., Φ 3  and Φ 4  corresponding to currents in legs  3  and  4  are coupled to low-reluctance center post  1203 - 3 ). As a result, currents in secondary windings  708 - 1  and  708 - 2  are each proportional to the respective current in primary windings  702 - 1  and  702 - 2  (i.e., the parallel current-doubler rectifiers  707 - 1  and  707 - 2  share load current I LOAD  equally). Otherwise, i.e., when fluxes Φ 3  and Φ 4  in legs  3  and  4  are coupled, the currents in secondary windings  708 - 1  and  708 - 2  are not equal, even though the primary currents in  702 - 1  and  702 - 2  are the same, due to the internal impedance of each secondary circuit. 
     The flux in low-reluctance center post  1203 - 3 , which is shown in FIG. 13 as being equal to the sum of the fluxes of legs  1 - 4 , can be reduced by having opposite winding orientations in the windings of transformers in legs  3  and  4 , and in the filter-inductor legs  1  and  2 . FIG. 14 shows such a configuration in converter  1400 . (Note the difference between the dot positions of the windings in FIGS. 12 and 14.) With opposite winding orientations, both fluxes Φ 3  and Φ 4  and fluxes Φ 1  and Φ 2  flow in opposite directions through center post  1203 - 3 . As a result, the total flux Φ c  in un-gapped center post  1203 - 3  is reduced, thus relieving reducing the area in center post  1203 - 3  necessary to prevent saturation. 
     The integrated magnetic approach in FIGS. 10,  11 ,  12 , and  14  can be applied to any number of rectifier stages, although the integrated magnetic components in FIGS. 12 and 14 may require custom-designed magnetic cores when more than two parallel rectifier stages are present, because each additional rectifier stage requires an additional leg. For an even number of rectifier stages, the converter can be implemented with a number of x-type cores, using an x-core to integrate each pair of rectifiers, as illustrated by converters  1200  and  1400  of FIGS. 12 and 14. Finally, converters  700 ,  900 ,  1000 ,  1100 ,  1200 , and  1400  of FIGS. 7,  9 ,  10 ,  11 ,  12 , and  14  can be implemented using synchronous rectifiers, rather than diode rectifiers. 
     The current-sharing performance of each of converters  700 ,  900  and  1000  was verified experimentally on a 200 kHz, 100 A/2.5 V prototype designed to operate from a 48-volt input. The prototype was implemented with a half-bridge inverter and two current-doubler rectifier stages. The measured full-load current-sharing performance and conversion efficiency are summarized in Table I. 
     
       
         
               
             
               
               
               
               
             
               
               
               
               
             
           
               
                 TABLE I 
               
             
             
               
                   
               
               
                 Measured current-sharing performance and 
               
               
                 conversion efficiency of a 100-A/5-V prototype with 
               
               
                 two paralleled rectifier stages 
               
             
          
           
               
                   
                 First rectifier 
                 Second rectifier 
                   
               
               
                   
                 (i.e., rectifier 
                 (i.e., rectifier 
                   
               
               
                   
                 707-1) output 
                 707-2) output 
                   
               
               
                 Implementation 
                 current (A) 
                 current (A) 
                 Efficiency (%) 
               
               
                   
               
             
          
           
               
                 Non-coupled 
                 48.1 
                 48.6 
                 73.7 
               
               
                 inductors (e.g., 
               
               
                 converter 700 of 
               
               
                 FIG. 7) 
               
               
                 Coupled 
                 48.7 
                 47.8 
                 73.7 
               
               
                 inductors (e.g., 
               
               
                 converter 900 of 
               
               
                 FIG. 9) 
               
               
                 Integrated 
                 49.3 
                 48.1 
                 73.6 
               
               
                 Magnetics (e.g., 
               
               
                 converter 1000 
               
               
                 of FIG. 10) 
               
               
                   
               
             
          
         
       
     
     The detailed description above is provided to illustrate specific embodiments of the present invention and is not intended to be limiting. Numerous variations and modifications within the scope of the present invention are possible. The present invention is set forth in the following claims.