Abstract:
A broadband transmission line impedance transformer performs impedance transformation with improved frequency response and efficiency across a wide operational bandwidth. In particular, the bandwidth of a transmission line 2:1 impedance transformer may be significantly increased by adding an additional compensating capacitor as an internal component between interconnected transmission lines. This capacitor effectively improves low frequency response for a given length of transmission lines and decreases mismatch in an entire frequency range. The overall bandwidth ratio increases at least twice and mismatch decreases.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is a divisional of Ser. No. 11/224,972 filed Sep. 14, 2005 now U.S. Pat. No. 7,583,160, which claims priority to U.S. Provisional Application No. 60/610,692 filed on Sep. 17, 2004 entitled “Broadband Transmission Line Transformer” by Simon Y. London. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     This invention relates generally to broadband radio-frequency impedance transformers. More particularly, the invention relates to broadband transmission line transformers with non-integer turns ratio (fractional ratio transformers) and mostly for high power application. 
     2. General Description of the Prior Art 
     A particular class of RF impedance transformers with maximum achievable bandwidth and low insertion losses is a class of transmission line transformers that plays an important role in various RF systems, from low power up to high power levels. 
     The main frequency limitation factors of these transformers are shunt inductance at lower frequencies and electrical length of transmission lines at higher frequencies. These two contradictory factors determine the achievable bandwidth of transformers. Impedance transformers with diverse circuit models, having different interconnections of transmission lines and impedance transformation ratios, have different limitations influenced by these two factors. As result, greater or lower bandwidth can be achieved. 
     Widely used impedance transformation ratios are 4:1, 9:1 and 16:1 (integer turns ratios), and 1.5:1, 2:1, 3:1 (fractional turn&#39;s ratios). The latter ones are more difficult to realize at wide bandwidths, especially for high power application. 
     Various circuit diagrams of transmission line impedance transformers are presented in book of Jerry Sevick “Transmission Line Transformers.” Among the impedance transformers with non-integer turns ratios, the most necessary is 2:1 impedance transformer. A typical structure includes a two-way power combiner/divider, which consists of a combiner/divider itself and a 2:1 impedance transformer. 
     All of these RF transformers have multi-octave bandwidth and use generally ferrite toroids or other ferrite configurations. Due to high magnetic permeability of ferrite transformers, shunt inductance is high enough and it is possible to realize multi-octave bandwidth with admissible electrical length of transmission lines. 
     In high power transformers (5-100 kW), magnetic flux in ferrite is also high and introduces harmonics and intermodulation products. 
     Furthermore, for these transformers, hysteresis losses (heat dissipation) limiting power handling capability may require a liquid cooling system. Such transformers are heavy, expensive and can not be used in certain environmental conditions. 
     Many attempts to develop high power, broadband transformers without ferrite have been made. In this case the high-pass correction usually used for partly compensation of relatively small shunt inductance. In simplest case it may be one series connected capacitor at the input or at the output of transformer. 
     In spite of this, for achieving multi-octave bandwidth, especially at high power, the electrical length of the transformer&#39;s transmission lines should be great and high frequency limitation occurs. Additional low-pass correction compensates this effect to some extent. All of these corrections make transformers more complicated and expensive. 
     In addition, for transformers with fractional turn ratios, the impedance ratios in some practical cases are not close enough to integer numbers and, consequently, even if the transformer is ideal some mismatch occurs. For example, when typical turns ratio is 3/2, the corresponding impedance&#39;s ratio is 2.25, and with respect to required impedance transformation ratio equal to the calculated VSWR=1.125. Practically, in this case value of VSWR will be higher. 
     Furthermore, in combining the power of several amplifiers, a two-or more stage combining system is usually used. If each stage inserts some particular VSWR, the overall VSWR in the worst case is a product of its individual values. To decrease the above-mentioned theoretical value, the turns ratio 7/5 instead of 3/2 may be used, for example. A corresponding transformer is too complicated, especially for high power application. Besides, admissible electrical length of its transmission lines should be relatively small and the highest operating frequency decreases. 
     The factors discussed above are applicable to impedance transformers that are unbalanced, balanced and baluns (balanced-to-unbalanced). Frequently it is difficult to provide good balance for high power, broadband baluns, especially for fractional turns ratios. 
     In a prior art balun with 2.25:1 impedance ratio (U.S. Pat. No. 5,767,754), an additional transformer winding is used to improve balance. This winding introduces capacitive shunt effect that increases mismatch. Besides, balance can not be perfect due principally to the asymmetry of circuit models and the influence of stray elements, especially for high power applications. Different longitudinal voltages on the windings also introduce additional difficulties at high power levels. 
     Another approach is a chain connection of two transformers in different combinations (see books of Jerry Sevick: “Transmission Line Transformers” and “Building and Using Baluns and Ununs,” CQ Communications Inc., 1994). This approach is too complicate at high power levels and the balance is not good enough due to stray elements in real design. 
     In view of the above, it is an object of the present invention to provide a more effective, high power broadband impedance transformer. 
     It is another object of the present invention to provide a high frequency, high power transformer with unbalanced ports that is simple in construction and has a wide bandwidth without ferrite. 
     It is still a further object of the present invention to provide an unbalanced impedance transformer without ferrite having a multi-octave bandwidth ratio up to 20:1. 
     Still another object of the present invention is to provide a high power, broadband unbalanced transformer with a fractional turns ratio, and specifically to provide a 2:1 impedance transformation ratio. 
     Yet another object of the present invention is to provide a broadband, unbalanced transformer with a simple correction. 
     It is still a future object of the present invention to provide a broadband, unbalanced impedance transformer having very small mismatch with respect to standard nominal port impedances. 
     It is another object of the present invention to provide a high frequency, high power transformer with balanced ports that is simple in construction and has wide bandwidth without ferrite. 
     It is still a future object of the present invention to provide a broadband balanced-to-unbalanced impedance transformer (balun) having all above mentioned properties and good balance in entire frequency band. 
     SUMMARY OF THE INVENTION 
     According to the present invention, a significant increase in bandwidth and a simplifying, multi-octave impedance transformer are achieved. These results are obtained by combining two factors in one device:
         High admissible electrical length of transmission lines in a simple schematic model; and   usage of a correcting capacitor as an internal component between interconnected transmission lines.       

     This capacitor, together with shunt inductance of transmission lines, effectively decreases mismatch in the entire frequency band caused by 3/2 turn&#39;s ratio. 
     The described effect takes place for unbalanced-to-unbalanced transformers, for balanced-to-balanced transformers and for balanced-to-unbalanced transformers (baluns). 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
       The above described features and advantages of the present invention will be more fully appreciated with reference to the detailed description and figures, in which: 
         FIG. 1  illustrates the block diagram of a typical usage of a broadband impedance transformer having a preferable 2:1 impedance transformation ratio and incorporated with two-way power combiner/divider according to the prior art. 
         FIG. 2  illustrates a 2.25:1 broadband impedance transformer constructed with coaxial cables according to the prior art. 
         FIG. 3  illustrates a 2.25:1 broadband impedance transformer that consists of three-conductor transmission line according to the prior art. 
         FIG. 4  illustrates a 2.25:1 ratio impedance transformer that consists of three matched transmission lines, and specifically coax cables according to the prior art. 
         FIG. 5  illustrates a 2.25:1 ratio impedance transformer that consists of coaxial cables with identical characteristic impedances according to the prior art. 
         FIG. 6  illustrates a 2.25:1 impedance ratio balanced-to-balanced impedance transformer according to the prior art. 
         FIG. 7  illustrates the block diagram of a broadband impedance transformer with lumped correction elements according to the prior art. 
         FIG. 8A  illustrates 2:1 impedance ratio unbalanced transformer according to an embodiment of the present invention. 
         FIG. 8B  illustrates the version of  FIG. 8A  that consists of three-conductor line according to an embodiment of the present invention. 
         FIG. 9A  illustrates 2:1 impedance ratio balanced transformer according to an embodiment of the present invention. 
         FIG. 9B  illustrates the version of  FIG. 9A  that includes two identical three-conductor lines according to an embodiment of the present invention. 
         FIG. 10  illustrates a balun transformer according to an embodiment of the present invention. 
         FIG. 11  illustrates a balun transformer with correcting capacitors according to an embodiment of the present invention. 
         FIGS. 12   a,b  illustrate an experimental VSWR characteristic of a two-way power combiner incorporated into a transformer according to an embodiment of the present invention. 
         FIG. 13  is a graph of experimental insertion loss characteristics a of two-way power combiner incorporated into a transformer according to an embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION 
     Referring to  FIG. 1 , there is typical prior art arrangement  1  when a 2:1 impedance ratio transformer  2  is required. Widely used broadband power combiners/dividers  3  have, at common output/input port  4 , the parallel connection of two 50-Ohm transmission lines. Inside combiner/divider these lines (or frequently coaxial cables) may be interconnected in various ways, depending on the schematic of the device, but two inputs/outputs  5  and  6  still have nominal 50-Ohm impedance. By a 2:1 impedance ratio transformer  2 , the nominal impedance at port  7  will be also 50 Ohm. 
     At high power and in a broadband application, where efficiency is an important factor, transmission line impedance transformers are the best in most cases of HF-VHF frequency bands. These transformers generally have a simple construction. 
     Referring to  FIG. 2 , there is electrical scheme of one of such transformer  10 , investigated in above-mentioned book of Jerry Sevick. This transformer consists of paired coax cables  14  and  17  with their inner conductors  15  and  18  correspondingly. Paired outer conductors  16  and  19  form the second turn of transformer. Conductors  15  and  18  form first and third turns correspondingly. 
     The nominal impedance at port  11  with respect to common ground  13  is 2.25 times more than the nominal impedance at port  20  with respect to ground  13 . Consequently, this unbalanced transformer with fractional 3/2 turns ratio, even if ideal, implies VSWR=1.125. Shunt inductance increases this value at lower frequencies. 
     Besides, this transformer can operate satisfactorily if electrical length each of its transmission line does not exceed 60 deg at upper operating frequency. Corresponding optimum characteristic impedances of two coax cables  14  and  17  are different and non-standard values. For equal or standard values of characteristic impedances maximum admissible electrical length decreased rapidly. 
     Another electrical scheme of simple impedance transformer with the same impedance transformation ratio 2.25 and near the same achievable frequency characteristics is shown on  FIG. 3 . The spacing between adjacent conductors  23  and  24 , as well as spacing between adjacent conductors  24  and  25  are critical parameters to obtain maximum high frequency response. Two ports  26  and  28  are unbalanced with respect to common ground  29 . The main distinction between transformers shown on  FIG. 2  and  FIG. 3  is a different mutual arrangement of conductors. 
     Referring to  FIG. 4 , there is an electrical schematic of another prior art 2.25:1 ratio unbalanced impedance transformer. It consists of three matched transmission lines  33 ,  34  and  35  having equal characteristic impedances. This transformer is described in the article of S. E. London and S. V. Thomashevich, “Line Transformers with Fractional Transformation Factor,” Telecommunication and Radio Engineering, vol. 28/29, April 1974, pp. 129-131 and in the book of Jerry Sevick “Building and Using Baluns and Ununs,” CQ Communications Inc., 1994). 
     Ideally, this transformer with unbalanced ports  31  and  32  with respect to common ground  36  is operable at an unlimited upper frequency. On the other hand, it consists of two separate shunt inductances, formed by outer conductors of lines  33  and  34 , and of three separate transmission lines. Implementation of this transformer in high power applications introduces stray inductances and capacitances that decrease the upper operating frequency. 
     Moreover, at some electrical length, all transmission lines have a resonance cut-off frequency that may occur. As a result, these transformers are relatively complicated and operate also at limited electrical length of transmission lines. 
     Another prior art transformer ( FIG. 5 ) is obtained from the transformer of  FIG. 4  if the length of line  35  equals zero, and if two outer conductors of lines  33  and  34  are connected together at their equi-potential points. These lines can be paired as shown on  FIG. 5 . 
     This 2.25:1 ratio impedance transformer with two unbalanced ports  51  and  52  with respect to common ground  53  has the same characteristic impedance of both lines  54  and  57 . The line  54  with inner conductor  55  and outer conductor  56  corresponds to line  32  on  FIG. 4 . The line  57  with inner conductor  58  and outer conductor  59  corresponds to line  36 . Line  35  on  FIG. 4  is excluded. This transformer has features with respect to the transformers of  FIG. 2  and  FIG. 3  in mutual arrangement of conductors. This mutual arrangement provides satisfactory operation up to electrical length of each line 105 deg. (as described in the article in “Telecomm. and Radio Eng.”, 1974). Besides, the optimum characteristic impedances of lines  54  and  57  are equal and the same as transformer  FIG. 4 . 
     Referring to  FIG. 6 , there is a prior art electrical schematic of a 2.25 ratio balanced to balanced impedance transformer  60 , which has practically the same frequency limitations as the transformer shown on  FIG. 5 . The nominal impedance at balanced port  61 - 61 ′ is 2.25 times more than the nominal impedance at balanced port  62 - 62 ′. This transformer is symmetrical with respect to ground  63 . Two paired coax cables  64  and  65  are the same as cables  66  and  67 . Characteristic impedances of coax  64  and coax  66  are equal and two times less than characteristic impedances of coax cables  65  and  67 . 
     All transformers shown on  FIGS. 2-6  have low frequency limitations due to shunt inductances, which may be partly compensated (included in high-pass filter) by using additional components. 
     Referring to  FIG. 7 , there is a prior art block diagram of a broadband impedance transformer  70 , having unbalanced ports  73  and  74  with respect to common ground  77 . Compensating elements  72 ,  75  and  76  are connected typically at the input and at the output of transformer  70 . Capacitor  72  provides lower frequency correction; it forms high-pass filter with the transformer&#39;s shunt inductance  71 . Inductance  76  and capacitor  75  provides high frequency correction (see U.S. Pat. No. 5,309,120). 
     With this three-element correction, the transformers in U.S. Pat. No. 5,309,120 provide bandwidth ratio up to 5:1. They can operate satisfactorily at electrical length of lines significant less than 90 deg. 
     Referring now to  FIG. 8A , there is an electrical schematic of a 2:1 ratio impedance transformer  80  in accordance with the present invention. In this transformer having two unbalanced ports  81  and  82  with respect to common ground  90 , internal capacitor  83  plays two roles: 
     Effectively compensates shunt inductance of paired outer conductors  86  and  89 , and 
     Decreases inserted mismatch due to 3/2 turns ratio in a wide frequency band. 
     The optimum characteristic impedance of each of the coax cables  84  and  87  is equal Z 0✓ 2, where Z 0  is nominal impedance at port  82  (lower impedance side). 
     For transformers with a typical required 50:25 Ohm impedance transformation, the characteristic impedance of each coax, Z=35.35 Ohm, i.e., is practically 35 Ohm. Manufactured coax cable UT 141-35 has Z=35 Ohm. 
     Capacitor  83  in this transformer is connected between the end of inner conductor  85  of the first line  84  and port  82 . On the other hand, this capacitor is connected inside the transformer and between the first turn  85  and the second turn  88 . The third turn is formed by connecting together outer conductors  86  and  89  of coax cables  84  and  87 . 
     Capacitor  83 , together with the inductance of paired outer conductors  86  and  89 , forms a high-pass filter that also improves frequency response. As a result, this transformer has the following advantages: 
     Simple in construction (includes paired coax that have equal characteristic impedances), 
     Includes only one correcting element, 
     Operates satisfactorily up to electrical length of each coax 110 deg, and 
     Provides low reflection by relatively low shunt inductance. 
     The calculated value of reflection coefficient is ISImax 0.035 in cases of a 2:1 impedance transformation ratio. 
     Referring to  FIG. 8B , there is an electrical schematic of a 2:1 impedance transformer  91  according to the present invention, which is different from that shown in the  FIG. 8A  implementation of transmission lines. Instead of paired identical coax, there is a symmetrical three-conductor line with conductors  92 - 1 ,  92 - 2  and  92 - 3 . The capacitor  93  plays the same role as in the transformer, according to  FIG. 8A . 
     Nominal impedances at ports  94  and  95  with respect to common ground  96  are also the same as for  FIG. 8A . Therefore, the optimum characteristic impedance of the line formed by adjacent conductors  92 - 1  and  92 - 2  is the same as the characteristic impedance of line  84  in  FIG. 8A . The optimum characteristic impedance of the line formed by adjacent conductors  92 - 2  and  92 - 3  is the same as the characteristic impedance of line  87  on  FIG. 8A . In some practical cases this implementation of conductors is preferable for fabrication. 
     Referring to  FIG. 9A , there is an electrical schematic of a balanced-to-balanced 2:1 impedance transformer  100  according to an embodiment of the present invention. The nominal impedance at balanced port  101 - 101 ′ is twice more than nominal impedance at balanced port  102 - 102 ′. This transformer is symmetrical with respect to ground  109 . Paired coax cables  103  and  104  have the same characteristic impedances as cables  105  and  106  correspondingly. Characteristic impedances of coax  103  and coax  105  are equal and two times less than characteristic impedances of coax cables  104  and  106 . 
     The optimum characteristic impedance of each coax cable  103  and  105  is equal to Z/42, where Z is the nominal impedance at balanced port  102 - 102 ′ (lower impedance side). For a transformer with 100:50 Ohm impedance, the transformation characteristic impedance of each coax is equal Z=35.35 Ohm, i.e., practically 35 Ohm. 
     Two capacitors  107  and  108  have identical values of capacitances. They compensate shunt inductance of two pairs of outer conductors of coax cables  103 - 104  and  105 - 106 . The calculated reflection coefficient with these capacitors and with relatively small shunt inductance is 0.03 in the case of a 2:1 impedance transformation ratio. 
     Referring to  FIG. 9B , there is an electrical schematic of a 2:1 impedance transformer  110  in accordance with the present invention. This transformer is different from that shown on  FIG. 9A  implementation of transmission lines. Instead of paired identical coax cables, there are two symmetrical three-conductor lines with conductors  111 - 1 ,  111 - 2 ,  111 - 3  and  112 - 1 ,  112 - 2 ,  112 - 3  correspondingly. The capacitors  113  and  114  play the same role as capacitors  107  and  108  in the transformer, according to  FIG. 9A . Nominal impedances at balanced ports  115 - 115 ′ and  116116 ′ with respect to common ground  117  are also the same as for transformer shown on  FIG. 9A . 
     Now referring to  FIG. 10 , there is an electrical schematic of a 2.25:1 impedance ratio balun  210  according to an embodiment of the present invention. It consists of coax  211  that plays two roles. Its outer conductor (external surface) and conductors  212 ,  213 ,  214  and  215  form a balanced transformer with ports  218 - 218 ′ and  219 - 219 ′. The inner conductor and internal surface of the outer conductor (normally coax cable function) provide a balanced-to-unbalanced transition and form an unbalanced port  217 . This impedance transforming balun may be considered a result of an internal chain connection of simplest 1:1 balun and balanced-to-balanced impedance transformer (see S. London and S. Thomachevich, Pat. USSR, no 649050, 1979). Due to this internal chain connection of two transformers, the overall design is simpler than direct chain connection, and balance is better. These two factors are especially important for high power applications. The mutual arrangement of conductors in scheme  FIG. 10  is different with respect to that used in a balun according to Pat. USSR no. 649050. 
     Now referring to  FIG. 11 , there is an electrical schematic of a 2:1 impedance ratio transformer  310  accordance to an embodiment of the present invention. Coax cable  311  and conductors  312 ,  313 ,  314  and  315  operate exactly as coax cable  211  and conductors  212 - 215  in a balun transformer of  FIG. 10  correspondingly. Only additional capacitors  320  and  321  introduce the difference with respect to the balun transformer of  FIG. 10 . These two capacitors operate exactly as in balanced transformer shown on  FIG. 9B , and electrical characteristics are the same as for the balanced transformers of  FIG. 9A  and  FIG. 9B . 
     Experimental 20:1 Bandwidth Ratio Transformer 
     The laboratory prototype of an 50:25 Ohm impedance transformer was constructed without ferrite in accordance to  FIG. 9A  of present invention. It has been incorporated with two-way power combiner/divider as shown on  FIG. 1 , because it is the main application of such transformer. Besides, it verifies the possibility of designing a full device. Each of paired coax  84  and  87  on  FIG. 8  was produced from standard high power 50-Ohm coax FE  81  (15 kW @ f=−500 MHz). To obtain a characteristic impedance equal 35 Ohm, three upper layers of PTFE tape were removed. The transformer consists of three turns of paired these coax cable with average diameter 13.5 cm. Capacitor  83  shown on  FIG. 8  is formed as a parallel connection of six standard capacitors HEC HT-50 of 700 pF each. 
     A two-way power combiner consists of two cables FE  81  connected in parallel at common port  4  ( FIG. 1 ) that gives nominal impedance 25 Ohm this port. Experimental graphs are shown on  FIG. 10  and  FIG. 11 . As we can see on  FIG. 12 , the obtained VSWR max  in an operating frequency band from 2 to 40 MHz is close to a calculated value VSWR max =(1+(S |max)/(1−|S|max)=1.074, when |S| . . .  x  is equal ≈0.035, as pointed above. 
     The calculated upper operating frequency is equal z-, 43.5 MHz, i.e. enough close to an experimental result for a full device (transformer with combiner itself). Data on  FIG. 13  showing that full insertion losses of transformer and combiner are low verifies the practical importance of embodiments of the present invention. 
     While the devices and methods of this invention have been described in terms of specific embodiments, it will be apparent to those of skill in the art that variations may be applied to the devices without departing from the concept, spirit, and scope of the invention. Therefore, all such substitutions and modifications apparent to those skilled in the art are deemed to be within the spirit, scope, and concept of the invention as defined by the appended claims.