Abstract:
Kelvin (4-wire) connecting cables are routinely used when performing dynamic measurements (i.e., measurements with time-varying signals) on electrochemical cells and batteries. Current-carrying and voltage-sensing conductor pairs within such cables comprise distributed-parameter two-wire transmission lines which may extend several meters in length. As with all such transmission lines, internally reflected waves can oscillate back and forth at high frequency (hf) whenever the lines are not terminated in their characteristic impedances. Such hf reflected waves, by interacting with measuring circuitry, can seriously degrade low-frequency measurement accuracy. Apparatus is disclosed herein that suppresses hf reflected waves oscillating on Kelvin connecting cables during dynamic measurements of cells and batteries.

Description:
BACKGROUND OF THE INVENTION 
       [0001]    Electrical parameters measured with time-varying signals are referred to as dynamic parameters. The present invention relates to measuring dynamic parameters of electrochemical cells and batteries through Kelvin (4-wire) connecting cables. More specifically, it relates to suppressing high-frequency (hf) waves oscillating back and forth on a Kelvin cable&#39;s current-carrying and/or voltage-sensing conductors. 
         [0002]    Measuring automotive and standby cell/battery parameters with time-varying signals (i.e., measuring dynamic parameters) are now commonly accepted maintenance and diagnostic procedures. (See, e.g., U.S. Pat. Nos. 5,140,269, 6,262,563, 6,534,993, and 6,623,314). Because of the very small impedances of such cells/batteries, Kelvin (4-point) connections are routinely employed to reduce the influence of the contact and lead-wire resistances. Kelvin connections couple to each cell/battery terminal at two separate contact points—one for current and one for voltage. Apparatus for measuring a two-terminal cell/battery by means of Kelvin connections therefore requires a four-wire interconnecting cable. 
         [0003]    When using Kelvin cables with time-varying signals, distributed mutual-inductance between current-carrying and voltage-sensing conductors has been a problem. As disclosed in U.S. Pat. Nos. 7,106,070 and 7,425,833, mutual-inductance can be reduced by inserting a special cable section in tandem with the original Kelvin cable. This special section transposes conductors thereby introducing a negative mutual-inductance section to cancel the positive mutual-inductance of the original Kelvin cable. 
         [0004]    However, even after canceling a cable&#39;s mutual-inductance, a significant problem remains. The current-carrying conductors and the voltage-sensing conductors comprise two twisted-pair distributed-parameter transmission lines—not unlike those found in Category-5 Ethernet cables. These lines may extend over several meters in length. As with all distributed-parameter transmission lines, internal wave reflections can occur unless the lines are terminated in their characteristic impedances—a situation which virtually never occurs in practice. Such hf waves, oscillating back and forth, can interact with measuring circuitry to seriously degrade the accuracy of low-frequency dynamic measurements performed with circuitry connected through the Kelvin cables. Ironically, the very technique for reducing mutual-inductance described above introduces discontinuities that can actually contribute to such oscillations. Solving this previously-unrecognized wave-oscillation problem is the purpose of the present invention. 
         [0005]    Consider  FIG. 1 .  FIG. 1  depicts prior-art measuring circuitry  10  connected to cell/battery  20  by means of four-wire cable  30 , Y-junction  40 , and Kelvin conductors A, B, C, and D. Current-carrying conductors A and B couple to positive and negative cell/battery terminals at contact points  50  and  60 , respectively. Voltage-sensing conductors C and D separately couple to positive and negative cell/battery terminals at contact points  70  and  80 , respectively. During dynamic measurements, a time-varying current flows through current-carrying conductors A and B and also flows inside cell/battery  20  along an internal current path  90 . 
         [0006]      FIG. 2  shows a typical arrangement of conductors employed in prior-art apparatus such as that shown in  FIG. 1 . Measuring circuitry  10  comprises current-excitation circuitry  160 , voltage-sensing circuitry  170 , computation/control circuitry  180 , and display circuitry  190 . Current-excitation circuitry  160  and voltage-sensing circuitry  170  couple, respectively, to the A-B conductor-pair  140  of four-wire cable  30  at terminals  200  and  210 , and to the C-D conductor-pair  150  of four-wire cable  30  at terminals  220  and  230 . Computation/control circuitry  180  communicates bilaterally with both current-excitation circuitry  160  and voltage-sensing circuitry  170  and receives current- and voltage-signal inputs with which it computes dynamic parameters of cell/battery  20 . The results of this computation are communicated to the user through display  190 . 
         [0007]      FIG. 2  further discloses a spaced-apart cable section  35  comprising an A-C pair of insulated wires  120  contacting the positive terminal of cell/battery  20  at points  50  and  70 , respectively, and a B-D pair of insulated wires  130  contacting the negative cell/battery terminal at points  60  and  80 , respectively. Each of these conductor-pairs comprises a current-carrying conductor paired with a voltage-sensing conductor. Pairs  120  and  130  are necessarily spaced-apart at the cell/battery terminals but are brought into close proximity at Y-junction  40  where they are re-arranged for connection to four-wire cable section  30 . Throughout section  30 , the A-B current-carrying conductors and the C-D voltage-sensing conductors are separately paired and twisted together, pair  140  and pair  150 , respectively, to reduce mutual inductance between current-carrying and voltage-sensing circuits. The A-B and C-D conductors therefore comprise two twisted-pair distributed-parameter transmission lines of approximate length t. 
         [0008]      FIG. 3  shows current-excitation circuitry  160  of a type commonly employed in prior-art dynamic battery testing apparatus. Feedback excitation circuitry of this kind was first described by Wurst, et al., in U.S. Pat. No. 5,047,722. However, this early disclosure did not include Kelvin connections to the cell/battery, nor did it take into consideration the effect of the distance between the measuring circuitry and the cell/battery being tested. 
         [0009]    The A-B current-carrying conductors  360  of the battery-connecting cable are shown in  FIG. 3 . These conductors include twisted-pair  140  of section  30  as well as the A and B conductors of spaced-apart section  35  of  FIG. 2 . They may also include a mutual-inductance-canceling section, and their total length can extend several meters. 
         [0010]    The current-excitation circuitry  160  disclosed in  FIG. 3  comprises the series combination of resistor  300 , n-channel MOSFET  310 , and the A and B battery-cable terminals,  200  and  210 , respectively. This circuitry also includes operational amplifier  320  having its output terminal coupled to the gate of MOSFET  310  through resistor  350 . The common connection of resistor  300  and MOSFET  310  couples to the inverting (−) input of operational amplifier  320  through resistor  330 , thus providing negative feedback to amplifier  320 . As a result, the instantaneous voltage at the amplifier&#39;s inverting (−) input, R 300  x i(t), tracks the voltage v(t) applied to its non-inverting (+) input. Accordingly, computation/control circuitry  180  controls the current waveform i(t) flowing through cell/battery  10  by applying an appropriate voltage signal v(t) to the noninverting (+) input of amplifier  320 . Resistors  330 ,  350 , and capacitor  340  are compensation components—introduced specifically to ensure circuit stability at high frequency. 
         [0011]    Note that current i(t) can only pass through n-channel MOSFET  310  from drain to source. Accordingly, MOSFET  310  cuts off, and no current flows through cell/battery  20 , when v(t)&lt;0. Cell/battery current can only flow when v(t)&gt;0; and it can then only flow in the discharging direction. 
         [0012]    Similar feedback current-excitation circuitry, disclosed in U.S. Pat. Nos. 6,466,026 and 6,621,272, includes a p-channel MOSFET and a dc power supply. With that circuitry, v(t)&lt;0 causes the p-channel MOSFET to conduct—resulting in current flowing from the dc power supply into cell/battery  10  in the charging direction. Thus, cell/battery current can flow in either direction with the advanced circuitry disclosed in U.S. Pat. Nos. 6,466,026 and 6,621,272. In other respects, that circuitry functions just like the circuitry of  FIG. 3 . 
         [0013]      FIG. 4  shows a voltage waveform sometimes observed across series-resistor  300  in prior-art current-excitation circuitry  160  when it is exciting cell/battery  20  with a 22 Hz square wave. One notes large hf oscillations in the A-B current during conduction of MOSFET  310 . Close observations have shown that the frequency of these oscillations is greater than 10 MHz. Furthermore, the usual techniques for suppressing hf oscillations in feedback circuits, such as introducing compensation components  330 ,  340 , and  350 , or placing picofarad-size bypass capacitors at various points within the circuit, have proven to be surprisingly ineffective. Suppressing such oscillations is an object of the present invention. 
       SUMMARY OF THE INVENTION 
       [0014]    I have discovered that hf reflected waves on Kelvin cables can oscillate back and forth, thus causing seriously degraded low-frequency measurement accuracy. At high-frequency, the current-carrying conductors and the voltage-sensing conductors of Kelvin cables comprise two distributed-parameter twisted-pair transmission lines—not unlike those found in Category-5 Ethernet cables—which may extend several meters in length. As with all distributed-parameter transmission lines, internal wave reflections can occur unless the lines are terminated in their characteristic impedances—a situation which virtually never occurs in practice. Such oscillating reflected waves can interact with measuring circuitry to seriously degrade the accuracy of low-frequency dynamic measurements performed with circuitry connected through Kelvin cables. 
         [0015]    Apparatus for suppressing hf oscillations on Kelvin cables is disclosed herein. It comprises magnetic material surrounding the cable, and/or circuitry inserted at the input end and/or the output end of the cable&#39;s current-carrying and/or voltage-sensing conductors. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0016]      FIG. 1  is a drawing illustrating prior-art dynamic measuring circuitry connected to a cell/battery by means of Kelvin connections. 
           [0017]      FIG. 2  is a schematic representation of a prior-art conductor arrangement commonly employed with the apparatus of  FIG. 1 . 
           [0018]      FIG. 3  is a schematic diagram depicting prior-art current-excitation circuitry commonly employed in the dynamic battery testing apparatus of  FIG. 1 . 
           [0019]      FIG. 4  is an oscilloscope image showing the voltage waveform sometimes seen across series resistor  300  in a prior-art circuit such as that depicted in  FIG. 3 . 
           [0020]      FIG. 5  is a depiction of the A-B current-carrying conductors of  FIG. 2  modeled as a short-circuited transmission line. 
           [0021]      FIG. 6  is a frequency plot of the real part of the input impedance of the transmission line of  FIG. 5  near its quarter-wavelength resonance frequency. 
           [0022]      FIG. 7  is a frequency plot of the imaginary part of the input impedance of the transmission line of  FIG. 5  near its quarter-wavelength resonance frequency. 
           [0023]      FIG. 8  is a drawing of measuring apparatus which includes oscillation suppression devices  390 ,  400  and  410 . 
           [0024]      FIG. 9A  is a schematic diagram showing one embodiment of suppression circuitry  400  and/or  410  applied to either end of the current-carrying A-B transmission line. 
           [0025]      FIG. 9B  is a schematic diagram showing another embodiment of suppression circuitry  400  and/or  410  applied to either end of the current-carrying A-B transmission line. 
           [0026]      FIG. 9C  is a schematic diagram showing still another embodiment of suppression circuitry  400  and/or  410  applied to either end of the current-carrying A-B transmission line. 
           [0027]      FIG. 10A  is a schematic diagram showing one embodiment of suppression circuitry  400  and/or  410  applied to either end of the voltage-sensing C-D transmission line. 
           [0028]      FIG. 10B  is a schematic diagram showing another embodiment of suppression circuitry  400  applied to the input end of the voltage-sensing C-D transmission line. 
           [0029]      FIG. 10C  is a schematic diagram showing another embodiment of suppression circuitry  410  applied to the output end of the voltage-sensing C-D transmission line. 
           [0030]      FIG. 11  is an oscilloscope image showing the voltage waveform across resistor  300  after inserting suppression circuitry  400 —comprising a 10 μF bypass capacitor  420 —at the input of the A-B transmission line. 
           [0031]      FIG. 12  is a drawing of two cells connected in series showing the definitions of impedances ZA, ZB, and ZC measured in a 3-point impedance experiment. 
           [0032]      FIG. 13  is a frequency plot of the percent differences between (RA+RB) and RC for six series-connected cell-pairs measured in a 3-point impedance experiment before introducing any suppression circuitry. 
           [0033]      FIG. 14  is a frequency plot of the percent differences between (RA+RB) and RC for six series-connected cell-pairs measured in a 3-point impedance experiment after inserting suppression circuitry  400  comprising a 10 bypass capacitor  420  at the input end of the A-B transmission line. 
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0034]    Consider twisted pairs  140  and  150  of the 4-wire Kelvin cable section  30  depicted in  FIG. 2 . Except for the wire size, these twisted pairs are very much like the twisted-pair transmission lines found in Category-5 Ethernet cables. (See, e.g., http://en.wikipedia.org/wiki/Category — 5_cable). Accordingly, we will assume that these lines have characteristic impedances of Z 0 =100Ω and that their propagation velocities are v p =0.64·c, where c=3·10 8  m/s is the velocity of light in free space. We will also assume that the A-B transmission line has length l=2m and comprises twisted wires of size AWG # 12  (Cat-5 cables use AWG # 24 ). This assumption yields an approximate ac wire resistance of R ac =0.199 Ω/m near the quarter-wavelength frequency f λ/4 =v p /4l=24.0 MHz. (See, e.g., http://www.cvel.clemson.edu/emc/calculators/Resistance_Calculator/wire.html). 
         [0035]    The impedance of cell/battery  20  is typically only a few milliohms—a value much less than the A-B line&#39;s characteristic impedance Z 0 ≅100Ω. Accordingly, the cell/battery will be modeled herein as a short-circuit terminating the A-B transmission line. 
         [0036]    Consider  FIG. 5 . This figure depicts an A-B transmission line  140  of length l=2 m terminated in short-circuit  370 . According to well-known transmission line theory, the input impedance Z in  of such a transmission line can be written 
         [0000]        Z   in   =Z   0  tan  h (γ l )   (1)
 
         [0000]      where 
         [0000]      γ=α+ jβ   (2)
 
         [0000]      with 
         [0000]      α=2 R   ac   /Z   0    (3)
 
         [0000]      and 
         [0000]      β=2 πf/v   p .   (4)
 
         [0037]      FIGS. 6 and 7  show calculated frequency plots of the real and imaginary parts of Z in  in the vicinity of the quarter-wavelength frequency f λ/4 =24.0 MHz. One notes a very sharp resonance at 24.0 MHz—with the input resistance of the short-circuited A-B line rising above 12,000Ω at the resonance frequency. The calculated Q of this resonance is 
         [0000]        Q   λ/4   =πf   λ/4   /αv   p =99   (5)
 
         [0000]    which classifies this as a “high-Q” resonance. 
         [0038]    Accordingly, the A-B line&#39;s input impedance at frequency f λ/4  is very large compared with the excitation circuit&#39;s hf output impedance (R 300 ≅0.02Ω; R DS (on)≅0.005Ω). This ensures that any switching transients of MOSFET  310  that possess frequency components near f λ/4  will excite large hf voltage wavefronts on the A-B transmission line. Furthermore, the line&#39;s high Q, along with its unmatched terminations, ensures that such wavefronts, once excited, will undergo multiple reflections. These large oscillating wavefronts, if they arrive back at circuitry  160  in the proper phase, can cause MOSFET  310  to switch states—thus reinforcing this spurious excitation process. 
         [0039]      FIG. 8  discloses general means for suppressing such oscillations according to several embodiments of the present invention. One embodiment comprises magnetic core material  390  surrounding a section of four-wire cable  30  to introduce hf loss and attenuation into both two-wire transmission lines contained therein. Alternatively, suppression circuitry  400  or suppression circuitry  410  can be introduced at the cable&#39;s input interface between measuring circuitry  10  and four-wire cable  30 , or its output interface between four-wire cable  30  and spaced-apart cable section  35 . Suppression circuitry  410  may actually be an integral part of cable section  35  or of Y-junction  40 . 
         [0040]      FIGS. 9A ,  9 B, and  9 C disclose examples of suppression circuitry inserted at the input end  400  and/or the output end  410  of the A-B transmission line. This circuitry comprises bypass capacitor  420  connected across the A-B terminals to provide a hf signal-path between conductors, and/or magnetic cores  430  surrounding the A-B conductors to impede hf current flow. As shown in these three figures, the bypass capacitor and the magnetic cores can be either utilized separately, or in combination with one another. Such circuitry at the A-B line&#39;s input  400  functions by preventing excitation circuitry  160  from exciting spurious wavefronts on the A-B transmission line. Such circuitry at the line&#39;s output  410  prevents spurious wavefronts, once excited, from being reflected back. 
         [0041]    The C-D transmission line presents a different problem and must be treated differently. Unlike current excitation circuitry  160 , voltage sensing circuitry  170  cannot excite hf wavefronts on the line. Such wavefronts can, however, be excited by transient switching currents passing through cell/battery  20  during measurement. This mechanism can be particularly troublesome when measuring UPS and telecom cells/batteries while they are in service. 
         [0042]      FIG. 10A  discloses suppression circuitry similar to that disclosed in  FIG. 9B  applied to the C-D transmission line. This circuitry comprises bypass capacitor  425  along with magnetic cores  435 . Again, the bypass capacitor and the magnetic cores can be either utilized separately, or in combination with one another, and can be connected at the line&#39;s input end  400 , and/or its output end  410 . However, there is a significant difference between such suppression circuitry applied to the C-D transmission line and that applied to the A-B transmission line. The very large input impedance of voltage-sensing circuitry  170  compared with the very small output impedance of current-excitation circuitry  160  dictates that the value of bypass capacitor  425  connected across C-D conductors will be much smaller than that of bypass capacitor  420  connected across the A-B conductors. 
         [0043]      FIG. 10B  discloses another form of suppression circuitry that can be applied to the circuitry end  400  of the C-D transmission line. A resistance  440  of approximate value to the line&#39;s characteristic impedance—in series with blocking capacitor  450 —can be connected directly across the line. Blocking capacitor  450  is necessary to prevent the battery&#39;s dc current from flowing through resistance  440 . Because the hf input impedance of voltage-sensing circuitry  170  is much larger than resistance Z 0  in parallel, the C-D line will be essentially terminated in its characteristic impedance Z 0 —thus preventing hf reflections from occurring at the circuitry-end of the C-D transmission line. 
         [0044]      FIG. 10C  discloses a similar suppression technique that can be applied to the cell/battery-end  410  of the C-D transmission line. Resistances  460 , whose sum value approximates the line&#39;s characteristic impedance, are connected in series with the C and/or D conductors at the cell/battery-end. Resistances  460  may actually be an integral part of cable section  35  or of Y-junction  40 . Essentially no dc voltage drop occurs across these resistances because of the very small dc current flowing in the voltage-sensing circuit. Furthermore, because of the very small series impedance of cell/battery  20  (&lt;10 mΩ), the C-D line will be essentially terminated in its characteristic impedance Z 0 —thus preventing reflections from occurring at the cell/battery-end of the line. 
         [0045]      FIG. 11  is an oscilloscope image showing the voltage waveform across series resistor  300  after inserting suppression circuitry  400 , comprising bypass capacitor  420  depicted in  FIG. 9A , at the input of the A-B transmission line. The value of capacitor  420  is 10 μF. Such a surprisingly large value is necessary to completely suppress oscillations because of the very small hf output impedance of current-excitation circuitry  160  (R 300 ≅0.02Ω; R DS (on)≅0.005Ω). Note that the spurious high-frequency oscillations observed in prior-art  FIG. 4  have completely disappeared in  FIG. 11 . 
         [0046]      FIG. 12  is a drawing depicting a “3-point impedance measurement” experiment devised to investigate the effect of suppression circuitry upon measurement accuracy. Two cells, connected in series with a conventional strap, are open-circuited. The three impedances defined in FIG.  12 —ZA, ZB, and ZC—are then measured. If these three measurements are accurately performed, one should find that ZA+ZB=ZC—to within a high degree of accuracy. Any deviation from this simple result would indicate measurement error. 
         [0047]    This 3-point experiment was performed before, and after, connecting the 10 μF bypass capacitor  420  across the A and B terminals at the A-B transmission line input  400 . The subject battery was a 25 Ah 6-cell Hawker Cyclon battery—chosen because of its exposed cell-terminals and interconnecting straps. The six cells of the open-circuited battery were measured as six pairs, each pair comprising two adjacent cells. Three complex impedance measurements were performed on each adjacent cell-pair at frequencies of 2.58, 22.22, 44.44, and 90.91 Hz.  FIGS. 13 and 14  show the results of these measurements. 
         [0048]      FIGS. 13 and 14  are frequency plots of the percent differences between (RA+RB) and RC for the six adjacent cell-pairs. (R is the real part of measured impedance Z.)  FIG. 13  displays prior-art results obtained from data measured before connecting the 10 μF bypass capacitor  420  across the A and B terminals at  400 .  FIG. 14  displays new results obtained after connecting the 10 μF bypass capacitor  420  across the A and B terminals at  400 . 
         [0049]    One sees from  FIGS. 13 and 14  that the 10 μF bypass capacitor dramatically improves measurement accuracy. Before inserting capacitor  420 , the observed percent differences varied from nearly 4% to more than 10%. A trend for the errors to increase with frequency is very apparent. After inserting capacitor  420 , all percent differences are less than 0.8%,and most are less than 0.4%. In addition, the frequency dependence of the errors has disappeared. 
         [0050]    This completes the disclosure of my invention. The invention comprises a magnetic material surrounding the cable, and/or circuitry inserted at the input end and/or the output end of the cable&#39;s current-carrying and/or voltage-sensing conductors. A particular embodiment of the invention simply comprises a large bypass capacitor connected directly across the current-carrying conductors at the interface between the measuring circuitry&#39;s output and the Kelvin cable&#39;s input. Other embodiments include magnetic cores placed on the current-carrying and/or the voltage-sensing conductors and/or characteristic-impedance resistances terminating the voltage-sensing conductors. These embodiments represent simple, yet effective solutions to an important, but previously unrecognized problem. 
         [0051]    Although suppression circuitry has been disclosed inserted at the line&#39;s input, its output, or both, it could also be inserted internally to the line, at say, the terminus of a mutual-inductance cancellation section. These, and other variations, will be apparent to those skilled in the art and are intended to be covered by the appended claims.