Abstract:
The output slew rate of a differential transmission line driver ( 13 ) can be limited by suitably controlling signal slew rates ( 52 ) at the control inputs (neg, pos) of the drive switches (M 1 -M 4 ) that control current flow through the load impedance (Rload) of the driver.

Description:
FIELD OF THE INVENTION 
     The invention relates generally to data transmission systems and, more particularly, to differential transmission line drivers. 
     BACKGROUND OF THE INVENTION 
     FIG. 1 illustrates an example of a conventional low voltage differential driver for driving a differential data transmission line. The example of FIG. 1 is a current mode driver wherein the polarity of the differential output voltage, y-z, is generated by the direction of current flowing through a load resistor Rload of a drive stage  13 . The direction of the current flow is controlled by two sets of nMOS (see M 1 -M 4 ) switches in the drive stage  13 , which switches are driven by a predrive stage illustrated at  11 . In the illustrated example, if the input signal, in, received by the predrive stage  11  is high, then the signal pos is driven high by operation of predrive switches P 2  and M 6 , and the signal neg is driven low by operation of the predrive switches P 1  and M 5 . Consequently, the drive switches M 1  and M 4  are turned on and the drive switches M 2  and M 3  are turned off, so current flows through the load resistor Rload from node y to node z. When the input signal to predrive stage  11  is low, then the signal neg is driven high and the signal pos is driven low, so current flows through the load resistor from node z to node y. 
     It is well known that electromagnetic interference (EMI) and switching noise in data transmissions can be reduced by slowing down rise and fall times and thereby eliminating fast, noisy transitions. Reflections within a transmission line will interfere with the transmitted signal if the time for the reflection to return to the beginning of the transmission line exceeds the transition time. This reflection time is dependent on the length of the transmission line. 
     It is therefore desirable to provide for a transmission signal with a longer transition time, because a longer transition time would permit a longer length transmission line. 
     It can therefore be seen that slew rate control circuitry is often beneficial in data transmission systems, particularly in systems which have relatively long transmission lines. Although output slew rate control has been addressed with respect to the rise and fall times of single-ended buffers, such single-ended techniques do not address the unique characteristics of differential drivers such as illustrated in the example of FIG.  1 . 
     It is therefore desirable to provide output slew rate control for a differential transmission line driver. 
     The present invention recognizes that, in a current mode differential driver such as the example shown in FIG. 1, the operation of the set of switches M 1 , M 4  interacts with the operation of the set of switches M 2 , M 3  during transitions of the differential output. Accordingly, the present invention further recognizes that these interactions between the operations of the sets of drive switches should be taken into consideration when applying output slew rate control to a differential driver. In this regard, the invention recognizes that, for example, an external capacitor coupled between the outputs y and z of FIG. 1 will reduce the output slew rate of the differential driver without affecting the switching characteristics of the switches M 1 -M 4 . However, a capacitor as large as 30 pf may be required to produce transition times greater than 1 ns. The area needed for this capacitance can become disadvantageously large, particularly if a multichannel device is implemented. Furthermore, many conventional data transmission standards, for example LVDS, require the driver to have a low output capacitance. 
     It is therefore desirable to provide output slew rate control for a differential line driver without increasing the output capacitance of the driver. 
     According to the invention, the output slew rate of a differential driver can be limited by suitably controlling signal slew rates at the control inputs of the drive switches that control current flow through the load impedance of the differential driver. This advantageously reduces EMI and switching noise, and permits use of longer transmission lines, without increasing the output capacitance of the differential driver. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 illustrates a conventional example of a differential transmission line driver. 
     FIG. 2 illustrates exemplary embodiments of a predrive stage for a differential transmission line driver according to the invention. 
     FIG. 3 illustrates an exemplary embodiment of the voltage regulator of FIG.  2 . 
     FIG. 4 illustrates in tabular format exemplary differential output voltage transition times achieved by the invention for various combinations of supply voltage and operating temperature. 
     FIG. 5 illustrates exemplary operations which can be performed by the embodiments of FIG.  2 . 
    
    
     DETAILED DESCRIPTION 
     FIG. 2 illustrates exemplary embodiments of a predrive circuit stage according to the invention which can be used to drive a differential drive circuit stage such as the differential drive stage  13  of FIG.  1 . The embodiment of FIG. 2 includes slew rate control circuits embodied as RC circuits. These RC circuits control the slew rates of voltage transitions of the signals neg and pos, that are used to control the drive switches of the differential drive stage. For example neg would control the drive switches M 2  and M 3  of FIG. 1, and pos would control the drive switches M 1  and M 4  of FIG. 1, thereby providing the desired slew rate control at the differential output of FIG.  1 . 
     As shown in FIG. 2, the combination of resistor R 1  and capacitor C 1  controls the time required for predrive switch P 1  to charge the neg node from voltage node  21 , and the combination of resistor R 2  and capacitor C 2  controls the time required for the predrive switch P 2  to charge the pos node from voltage node  21 . Also, the combination of the resistor R 3  and capacitor C 1  controls the time required for the predrive switch M 5  to discharge the neg node to ground potential, and the combination of resistor R 4  and capacitor C 2  controls the time required for the predrive switch M 6  to discharge the pos node to ground potential. The above-described operations with respect to the neg node are permitted by the illustrated series connection of the switch P 1  and resistor R 1  between the voltage node  21  and the neg node, the series connection of the resistor R 3  and the switch M 5  between the neg node and ground, and the connection of capacitor C 1  between the neg node and ground. The above-described operations with respect to the pos node are permitted by the analogous connection of the elements P 2 , R 2 , R 4 , M 6  and C 2  with respect to the voltage node  21 , the pos node and ground. 
     The charging resistors R 1  and R 2  are advantageously much smaller than the discharging resistors R 3  and R 4 , in order to offset suitably the charging and discharging voltage characteristics at the neg and pos nodes. If the RC circuits were symmetrical, the voltage transitions at neg and pos would cross at exactly Vcc/2, which is typically 1.65 volts. This voltage is too low to turn on the drive switches (for example M 1 -M 4  of FIG. 1) of a typical differential drive stage. If the voltages at neg and pos cross at Vcc/2, then all four switches of the differential drive stage would be turned off at that crossing time, effectively disabling the driver. 
     Accordingly, and as mentioned above, the charging resistors R 1  and R 2  should typically be much smaller than the discharging resistors R 3  and R 4  such that, for example, the voltages at neg and pos cross at about 2.1 volts. At this crossing point, all four of the differential drive switches (e.g. M 1 -M 4  of FIG. 1) are conducting simultaneously, but the resistor values of R 1 -R 4  can be selected so that the operation of the set of switches driven by neg (for example M 2  and M 3  of FIG. 1) and the operation of the set of switches driven by pos (for example M 1  and M 4  of FIG. 1) do not substantially interfere with one another. In one exemplary embodiment, resistors R 1  and R 2  are approximately 500 ohms, resistors R 3  and R 4  are approximately 1800 ohms, and capacitors C 1  and C 2  are approximately 0.5 pf. 
     The predrive stage of FIG. 2 also includes a voltage regulator  23  which receives Vcc as input and produces a regulated output voltage Vr at node  21 . This regulator  23  is provided because, if the switches P 1  and P 2  were connected directly to Vcc as in FIG. 1, the RC operational characteristics would change greatly over the possible range of Vcc. In one example, the possible range of Vcc is 3.0 to 3.6 volts. Without the regulator  23 , the voltage Vcc would act as the initial voltage of the RC circuits of FIG. 2, thereby resulting in different charging/discharging rates for different values of Vcc. These different charging/discharging rates would cause correspondingly different transition times at the differential output, and would also cause the voltages at neg and pos to cross at correspondingly different points. Thus, changes in Vcc would disadvantageously affect the final output waveform of the differential drive stage. 
     The voltage regulator  23  is provided as a low dropout linear voltage regulator which produces the regulated voltage Vr. The regulated voltage Vr avoids undesirable variations in the transition times and waveforms produced at the output of the differential driver (for example across the load resistor Rload of FIG.  1 ). In one embodiment, the regulated voltage Vr is about 2.8 volts, for a Vcc range of 3.0 to 3.6 volts. 
     FIG. 3 illustrates an exemplary embodiment of the low dropout voltage regulator  23  of FIG.  2 . The negative input of an amplifier  30  is connected to a reference voltage of 1.2 V, which reference voltage can be generated by a bandgap circuit. The amplifier  30  regulates node  31  to 1.2V, which, via the two resistors, R 5  and R 6 , generates a voltage of 2.8V at the node Vr. A deviation from 2.8 V at node Vr will cause a corresponding deviation at node  31 , which will cause the output of the amplifier  30  to change in order to correct the error. For instance, when the driver (P 1 /P 2  in FIG. 2) switches, node Vr is transiently loaded because it must supply current for the predrive stage. Node Vr, and consequently node  31  are pulled down. This causes a decrease in voltage at the output of the amplifier  30 , node Cr, which increases the output current from P 3 . This current supplies some of the load current needed, and also eventually charges the node Vr back up to 2.8V. Capacitor C 3  is coupled between Vr and Cr to improve the response time of the regulator. When Vr is pulled down during transient loading, node Cr will also be immediately pulled down, allowing P 3  to begin supplying current before the amplifier kicks in to decrease the voltage at Cr. Capacitor C 4  is used as a charge storage device to provide loading current and to minimize the drop in node Vr when the driver (P 1 /P 2  in FIG. 2) switches. 
     FIG. 4 illustrates in tabular format exemplary rise times and fall times of a differential output signal (for example y-z in FIG. 1) produced by an example implementation of a differential transmission line driver according to the invention, at various Vcc values and various operating temperatures, using the aforementioned exemplary resistor values, using a conventional 0.6 micron BiCMOS process, with the resistors R 1 , R 2 , R 3  and R 4  having a temperature coefficient of 50 ppm, and using an input signal of 100 Mbps. As shown in FIG. 4, and assuming nominal process parameters, the rise times and fall times vary by only about 200 ps (+ or −5%) over Vcc and temperature. 
     FIG. 5 illustrates exemplary operations which can be performed by the predriver embodiment of FIG.  2 . At  51 , the predrive circuit is activated by a change in the level of its input signal (in/in′). At  52 , slew rate control is applied to an output of the predrive circuit in order to limit the slew rate of the predrive output (neg, pos). At  53 , the predrive output is used to control corresponding switches of the differential drive stage (for example, M 1 -M 4  of the differential drive stage shown at  13  in FIG.  1 ). 
     As demonstrated above, the present invention provides a relatively simple predrive circuit which can advantageously produce output transition times of a few nanoseconds that vary by an extremely small amount over Vcc and temperature. By maintaining a desired transition time within a specified margin, parameters such as minimum noise, stub length and maximum signaling lengths can advantageously be specified. 
     Although exemplary embodiments of the invention are described above in detail, this does not limit the scope of the invention, which can be practiced in a variety of embodiments.