Abstract:
An apparatus is described having a reference voltage circuit. The reference voltage circuit includes a diode to receive first and second currents having first and second respective current densities, where, the first and second current densities are different and determined by circuitry that precisely controls the respective amount of time the first and second currents flow into the diode. The reference voltage circuit also comprises circuitry to form a reference voltage by combining first and second voltages generated from respective voltages of the diode that result from the first and second currents flowing through the diode.

Description:
FIELD OF INVENTION 
       [0001]    The field of invention pertains generally to the semiconductor arts, and, more specifically, to a bandgap reference circuit with capacitive bias. 
       BACKGROUND 
       [0002]    Circuits often require a precise voltage level. As such, reference voltage circuits have been developed to generate a precise voltage level that, ideally, does not drift or largely vary with, e.g., temperature changes that the reference voltage circuit may be subjected to. One type of reference voltage circuit, referred to as a bandgap voltage reference circuit, generates a proportional to absolute temperature (PTAT) voltage and a proportional to complementary temperature (CTAT) voltage. The PTAT voltage is derived from a pair of voltages that are generated from different current densities through a P-N junction. The accuracy of the reference voltage that is output by the circuit is most sensitive to the PTAT voltage, since the PTAT voltage is usually multiplied by a certain factor (˜10) to achieve temperature-compensation at the reference voltage output. In turn, the PTAT voltage is proportional to the logarithm of the ratio of the two different current densities. 
         [0003]    A problem with integrating a bandgap reference voltage circuit onto a semiconductor chip manufactured with a logic manufacturing process is that the ratio of the two current densities is typically limited to one order of magnitude (e.g., no higher than 50). The relatively small current density ratio results in a bandgap reference circuit that is more sensitive to circuit non-idealities, like amplifier or device mismatch, and may therefore not be suitably accurate or stable for its particular application. Moreover, the ratio of current densities in prior art solutions depends on the matching of devices, like MOS transistors or resistors, which limits the achievable accuracy. In modern technologies it is further difficult to integrate analog structures like resistors, current sources or amplifiers, with sufficient performance (ideality). The requirements for a technology would be relaxed, if a reference circuit does not need such components, but can operate by similar means like the digital core circuitry. 
     
    
     
       FIGURES 
         [0004]    A better understanding of the present invention can be obtained from the following detailed description in conjunction with the following drawings, in which: 
           [0005]      FIGS. 1 a  and 1 b    show capacitive discharge behavior; 
           [0006]      FIGS. 2 a  through 2 d    show different phases of operation of a bandgap reference circuit having capacitive bias; 
           [0007]      FIG. 3  shows exemplary voltages of the four phases of  FIGS. 2 a    through  2   d;    
           [0008]      FIG. 4  shows an embodiment of a bandgap reference circuit having capacitive bias; 
           [0009]      FIGS. 5 a  through 5 e    show different switch states of the circuit of  FIG. 4 ; 
           [0010]      FIG. 6  shows exemplary waveforms of the circuit of  FIG. 4 ; 
           [0011]      FIG. 7  shows another bandgap reference circuit; 
           [0012]      FIG. 8  shows a computing system. 
       
    
    
     DETAILED DESCRIPTION 
       [0013]    A solution is to use “capacitive bias” in which first and second charged capacitors behave as the source of potential that drives first and second current densities through a P-N junction. The first and second current densities correspond to the pair of current densities whose ratio determines the PTAT voltage. The combination of P-N junction and capacitors operate in a region where each of their respective voltage decay behavior varies according to the natural log of time without any substantial dependence on (initial) biasing conditions or parasitics. 
         [0014]    As such, extremely large current density ratios can be achieved if one of the capacitors discharges for a small amount of time while the other of the capacitors discharges for a longer amount of time. Here, current density ratios of perhaps 4 or 5 orders of magnitude (e.g., 10,000) can be produced which, in turn, corresponds to a significantly larger generated PTAT voltage and a more accurate/stable reference voltage. Moreover, as described in more detail below, in various embodiments, the current densities can be accurately controlled by time, rather than by device sizing. 
         [0015]      FIGS. 1 a  and 1 b    show a simple capacitor discharge circuit whose behavior corresponds to the capacitive bias principle discussed just above. As shown in  FIG. 1   a,  initially, a switch SW is set in a first position that permits a supply voltage VDD to charge a capacitor C. After the capacitor is charged (e.g., approximately to the supply voltage), as shown in  FIG. 1B , the switch is closed in a second position to permit the charged capacitor to discharge into a P-N junction (diode D). 
         [0016]    It can be shown that the voltage across the diode D, which corresponds to the decaying voltage of the capacitor C, will behave as expressed in Eqn. 1 below: 
         [0000]        V   D ( t )=− V   T (ln(1−(1−exp(− V   DD   /V   T ))(exp(− I   S /( CV   T )) t )   Eqn. 1
 
         [0000]    Where V T  is the thermal potential (i.e., kT/q where k is Boltzman&#39;s constant, T is the temperature and q is elemental charge), V DD  is a supply voltage, I S  is the saturation current of the diode, C is the capacitance of the capacitor and t is the time where t=0 when the switch is closed to connect the capacitor C to the diode D. 
         [0017]    Of importance, Eqn. 1 can be approximated as three different equations depending on the value of t. That is, referring to equations  2   a,    2   b  and  2   c  below, a first equation  2   a  approximates Eqn. 1 for small values oft (e.g., less than 50 ns), a second equation  2   b  approximates Eqn. 1 for medium values oft (e.g., greater than 50 ns but less than 200 μs) and a third equation for large values of t (e.g, greater than 200 μs). 
         [0000]        V   D ( t )=− V   T ln((exp(− V   DD   /V   T )+((− I   S /( CV   T )) t ))   Eqn. 2a
 
         [0000]        V   D ( t )=− V   T ln(( I   S /( CV   T )) t )) Eqn. 2b
 
         [0000]        V   D ( t )=− V   T ln(1−exp(− I   S /( CV   T )) t ))   Eqn. 2c
 
         [0018]    Notably, the medium time approximation of Eqn. 2b varies as the natural log of time and does not depend on V DD . As such, a bandgap reference voltage circuit can be built that uses the relationship expressed in  FIG. 2 b    to precisely control the voltage across the PN junction, and, in so doing, precisely control the resulting current density through the PN junction. 
         [0019]    As explained in more detail below, the precise control is effected by precisely controlling the amount of time that the capacitor is allowed to discharge. Thus, with two capacitors that are precisely controlled in this manner, one for a shorter time and one for a longer time, a large dynamic range between the respective current densities that result from their discharge can be realized. A large dynamic range between these current densities corresponds to a large current density ratio, which, as described just above, can be used to generate a large PTAT voltage and corresponding accurate/stable output reference voltage. 
         [0020]      FIGS. 2 a  through 2 d    show the basic principle of operation of a capacitive bias based bandgap reference circuit. As observed in  FIG. 2 a   , during a first phase, a first circuit switching state connects a supply voltage V DD  to first and second capacitors C 1 , C 2  to charge them approximately to the supply voltage V DD . After the capacitors C 1  and C 2  are charged, the first phase ends and a second phase begins. Here,  FIG. 3  shows the behavior of the voltage on both capacitors C 1  and C 2 . As can be seen from  FIG. 3 , the voltages of both capacitors C 1  and C 2  charge together in lock stop during the first phase over time T 1 . 
         [0021]    As observed in  FIG. 2 b   , the second phase includes a second switching state in which capacitor C 2  is permitted to discharge through the diode D 1 . In an embodiment, the amount of time T 2  in which capacitor C 2  is permitted to discharge is tightly controlled and corresponds to a medium amount of time in which the discharge behave is largely defined by Equation 2b above. Here, as discussed at length above, Equation 2b describes decaying voltage behavior that varies as the natural log of the time that the capacitor is permitted to discharge. As such,  FIG. 3  shows the voltage of capacitor C 2  decaying according to natural log behavior for time T 2  during the second phase. When the tightly controlled time period T 2  that C 2  is permitted to discharge elapses, the second phase ends and the third phase begins. 
         [0022]    As observed in  FIG. 2 c   , the third phase includes a third switching state in which capacitor C 1  is permitted to discharge through the diode D 1 . In an embodiment, as with the discharge of capacitor C 2  during phase T 2 , the amount of time T 3  that capacitor C 1  is permitted to discharge is tightly controlled and is within a time period in which the behavior of the discharge is largely described by Eqn. 2b above. However, capacitor C 1  is permitted to discharge for a noticeably longer time period T 3  than capacitor C 2  is permitted to discharge T 2 . As such, as observed in  FIG. 3 , the voltage of capacitor C 1  decays to a noticeably lower voltage level VD 1  than the level VD 2  that capacitor C 2  decayed to. 
         [0023]    The difference in these voltage levels VD 1 , VD 2  corresponds to two significantly different current densities that flow through diode D 1  which, as described at length above, corresponds to a much higher current density ratio than is typical for a bandgap reference voltage circuit. For instance, current densities having ratios that are on the order of two orders of magnitude (100s), three orders of magnitude (1000s), four orders of magnitude (10000s) or even five orders of magnitude (100000s) are possible. 
         [0024]    As described in more detail below, in various embodiments, the second and third phases and their corresponding controlled capacitor discharge time periods T 2 , T 3  are designed to overlap rather than be disjointed as observed in  FIG. 3  (e.g., both phases start at the same time at the end of the first phase with the second phase ending noticeably before the third phase ends). 
         [0025]    After the amount of time T 3  that capacitor C 1  is permitted to discharge elapses, the third phase ends and a fourth phase begins. As observed in  FIG. 2 d   , in the fourth phase a fourth switching state is engaged in which the voltages of the two capacitors are effectively subtracted from one another and the voltage at the node between the two capacitors is expressed as: 
         [0000]        V _OUT=( C 2/( C 1+ C 2))( VD 2−(( C 1 /C 2) VD 1)   Eqn. 3
 
         [0000]    and where V_OUT further corresponds to the reference voltage generated by the circuit. Here, the four phases are continuously repeated to continuously generate the V_OUT reference voltage. 
         [0026]      FIG. 4  shows one embodiment of a circuit that is designed consistently with the principles discussed just above with respect to  FIGS. 2 a    through  2   d.    FIGS. 5 a    through  5   d  show the different switching states of the circuit of  FIG. 4 .  FIG. 6  shows the capacitor voltages. 
         [0027]    According to the operation of the circuit of  FIG. 4 , as observed in  FIG. 5 a   , in the first phase during time T 1  transistors P 1 , P 2  and N 1  are on while transistors P 0 , N 2 , N 3  and N 4  are off. With this switching arrangement, a direct current path from the supply voltage node to the ground references is established through both capacitors C 1 , C 2 . As a consequence, both of the capacitors C 1 , C 2  will be charged approximately to the supply voltage VDD. As such, as observed in  FIG. 6 , the voltage levels of both capacitors charge together up to the supply voltage within time T 1 . 
         [0028]    As alluded to above, in various embodiments, the discharge phases of the two capacitors may be overlapped such that they both start discharging at the same time but one of the capacitors stops discharging before the other of the capacitors stops discharging. The particular circuit of  FIG. 4  conforms to such an approach. As such, the second and third phases start at the same time at the end of the first phase. 
         [0029]    As observed in  FIG. 5 b   , the combined second and third phases (which takes place over time T 2 ) includes the discharging of the both capacitors C 1  and C 2  into diode D 1 . The discharge of capacitor C 2  is effected by turning transistors P 1  and P 2  off, keeping transistors N 3  and N 4  off and turning transistors P 0  and N 2  on. Additionally, the third phase, in which capacitor C 1  discharges into diode D 1 , is effected by keeping transistor N 1  on. With this switch state, both capacitors C 1  and C 2  are able to discharge into diode D 1 . Here, with both of capacitors C 1  and C 2  discharging together, referring to  FIG. 6 , note that the voltage of both capacitors fall together according to the logarithmic behavior of Eqn. 2b. 
         [0030]    After the shorter time period T 2  for discharging capacitor C 2  has elapsed, as observed in  FIG. 5 c   , transistor PO is turned off which prevents any further discharge from capacitor C 2  into diode D 1 . Here, referring to  FIG. 6 , C 2  is isolated so that it holds its voltage level VD 2 . The third phase continues and capacitor C 1  continues to discharge into diode D 1 . As such, its voltage continues to decay in accordance with Equation 2b. 
         [0031]    Eventually the time at which C 1  is to stop discharging T 3  is reached and, as observed in  FIG. 5 d   , transistor N 2  is turned off to prevent any further current flow through the diode D 1 . That is, the leg of the circuit that includes diode D 1  is open so that no current can flow through diode D 1 . Preventing any current flow through diode D 1  effectively stops capacitor C 1 &#39;s discharge into diode D 1 . As observed in  FIG. 6 , the voltage level of capacitor C 1  has fallen to a lower level VD 1  than the voltage level VD 2  that capacitor C 2  is holding at. 
         [0032]    With the time for the discharge of capacitor C 1  having elapsed, a fifth switch state is effected for the fourth phase in which the voltage on capacitor C 1  is subtracted from the voltage on capacitor C 2  to generate an output voltage that corresponds to Eqn. 3 at output node  401 ,  501 . Thus, as observed in  FIG. 5 e   , transistors N 3  and N 4  are turned on and transistor N 1  is turned off. This switching arrangement causes the current path depicted in  FIG. 5 e    to arise. 
         [0033]    It is pertinent to point out that various other implementations are possible to achieve a bandgap reference with capacitive bias (e.g., by using passive or active (amplifiers) multiplication). For example, as observed in the circuit of  FIG. 7 , the same technique can also operate with negative voltages and thereby realize a (reverse) bandgap reference circuit with nwell substrate diodes. Nwell substrate diodes are normally reverse biased, but more reliable and precise than PNP devices, due to the buried nature of the nwell. 
         [0034]    As another example, circuits having only one capacitor instead of two capacitors may be implemented. For example, the same principle works if a single capacitor is used, where the single capacitor is repeatedly re-charged, then discharged through a diode, and its remaining voltage “transferred” by a suitable charge-transfer mechanism (e.g. a switched-capacitor amplifier] to a summing node/component. 
         [0035]    Further still, in circuits having more than one capacitor, a series connection of the capacitors is not necessary to perform the intended summation of voltages VD 1  &amp; VD 2  (or scaled versions of the same). Here, a parallel combination of capacitors is possible (e.g.,  FIG. 5 e    shows a parallel combination of +VD 2  and −VD 1 ). Alternatively, summation through active circuitry, such as a switched-cap amplifier, can be used. Generally, regardless of the precise implementation, a “charge-transfer” is performed resulting in effective voltage addition (maybe including some scaling and therefore different to Eqn. 3), to reach the same basic function of temperature compensated output voltage 
         [0036]    Although the circuit diagrams discussed above do not depict the circuitry that generates the appropriate signals for switching transistors into an on or off state, it should be understood that such circuits exist and have not been shown purely for illustrative ease. 
         [0037]      FIG. 8  shows a depiction of an exemplary computing system  800  such as a personal computing system (e.g., desktop or laptop) or a mobile or handheld computing system such as a tablet device or smartphone, or, a larger computing system such as a server computing system. The computing system may contain a capacitive bias bandgap reference circuit as described above. 
         [0038]    As observed in  FIG. 8 , the basic computing system may include a central processing unit  801  (which may include, e.g., a plurality of general purpose processing cores and a main memory controller disposed on an applications processor or multi-core processor), system memory  802 , a display  803  (e.g., touchscreen, flat-panel), a local wired point-to-point link (e.g., USB) interface  804 , various network I/O functions  805  (such as an Ethernet interface and/or cellular modem subsystem), a wireless local area network (e.g., WiFi) interface  806 , a wireless point-to-point link (e.g., Bluetooth) interface  807  and a Global Positioning System interface  808 , various sensors  809 _ 1  through  809 _N (e.g., one or more of a gyroscope, an accelerometer, a magnetometer, a temperature sensor, a pressure sensor, a humidity sensor, etc.), a camera  810 , a battery  811 , a power management control unit  812 , a speaker and microphone  813  and an audio coder/decoder  814 . The above described capacitive bias bandgap reference circuit may be specifically used, for instance, to generate a reference voltage within a wireless interface circuit such as a cellular/RF interface circuit that is integrated on a system-on-chip having the processing cores and the memory controller. 
         [0039]    An applications processor or multi-core processor  850  may include one or more general purpose processing cores  815  within its CPU  801 , one or more graphical processing units  816 , a memory management function  817  (e.g., a memory controller) and an I/O control function  818 . 
         [0040]    The general purpose processing cores  815  typically execute the operating system and application software of the computing system. The graphics processing units  816  typically execute graphics intensive functions to, e.g., generate graphics information that is presented on the display  803 . The memory control function  817  interfaces with the system memory  802 . The system memory  802  may be a multi-level system memory having, e.g., an emerging three dimensional non-volatile memory technology at at least one of the levels. 
         [0041]    Each of the touchscreen display  803 , the communication interfaces  804 - 807 , the GPS interface  808 , the sensors  809 , the camera  810 , and the speaker/microphone codec  813 ,  814  all can be viewed as various forms of I/O (input and/or output) relative to the overall computing system including, where appropriate, an integrated peripheral device as well (e.g., the camera  810 ). Depending on implementation, various ones of these I/O components may be integrated on the applications processor/multi-core processor  850  or may be located off the die or outside the package of the applications processor/multi-core processor  850 . 
         [0042]    Embodiments of the invention may include various processes as set forth above. The processes may be embodied in machine-executable instructions. The instructions can be used to cause a general-purpose or special-purpose processor to perform certain processes. Alternatively, these processes may be performed by specific hardware components that contain hardwired logic for performing the processes, or by any combination of programmed computer components and custom hardware components. 
         [0043]    Elements of the present invention may also be provided as a machine-readable medium for storing the machine-executable instructions. The machine-readable medium may include, but is not limited to, floppy diskettes, optical disks, CD-ROMs, and magneto-optical disks, FLASH memory, ROMs, RAMs, EPROMs, EEPROMs, magnetic or optical cards, propagation media or other type of media/machine-readable medium suitable for storing electronic instructions. For example, the present invention may be downloaded as a computer program which may be transferred from a remote computer (e.g., a server) to a requesting computer (e.g., a client) by way of data signals embodied in a carrier wave or other propagation medium via a communication link (e.g., a modem or network connection). 
         [0044]    In the foregoing specification, the invention has been described with reference to specific exemplary embodiments thereof. It will, however, be evident that various modifications and changes may be made thereto without departing from the broader spirit and scope of the invention as set forth in the appended claims. The specification and drawings are, accordingly, to be regarded in an illustrative rather than a restrictive sense.