Abstract:
A surface acoustic wave (SAW) expander based transmitter and correlator based receiver comprises SAW devices that perform expander or correlator functions based on the types of signals inputted to the SAW devices. The SAW devices incorporate chirp with adaptive interference and programmable coding capabilities. The SAW devices and method of operating the devices allow the implementation of very low power radios that overcome problems with temperature drift, lithography constraints and interference and jamming suffered by prior art implementations.

Description:
BACKGROUND OF THE INVENTION 
     1. Related Application Data 
     This application is a continuation of U.S. application Ser. No. 11/089,630 filed on Mar. 23, 2005 which claims priority from U.S. Provisional Application No. 60/558,173 filed Mar. 26, 2004. 
     2. Field of the Invention 
     The invention relates generally to electronic circuits and, more particularly, to circuits for wireless communication. 
     3. Description of the Related Art 
     Conventional radios have been extensively implemented using superhetrodyne and direct conversion architectures constructed from circuit blocks such as mixers, amplifiers, and RF filters. There are many chipsets on the market using various process technologies, but they are usually active circuits requiring a power source. One alternate approach is to construct a radio using a passive surface acoustic wave (SAW) expander and correlator. 
       FIGS. 1 and 2  show a representative transmitter  10  and receiver  20 , respectively, using the SAW approach. The main disadvantage of SAW correlator and expander implementations in the prior art is that they are implemented at an intermediate frequency (IF) to allow a local oscillator (LO) ( 12 ,  22 ) to compensate for inherent temperature drift and tolerances of the SAW device ( 11 ,  21 ) as a closed loop system. Although the SAW device ( 11 ,  21 ) itself is passive in this implementation, other active circuits are still required at the front end and for the LO and up/down converters, as shown. Without such compensation, the correlation performance and sidelobe levels become unacceptable over a practical temperature range, making the radio performance unacceptable. Historically, correlators/expanders have not been implemented directly at the radio frequency (RF) front end because of the temperature drift problem described above, but also because of problems with device to device manufacturing variance and difficulties with small feature size lithography required for operation above about 800 MHz. 
     Thus, it would be desirable to have a correlator/expander implemented at the RF front end if the issues of temperature variation and tight lithographic technology could be resolved. 
     SUMMARY OF THE INVENTION 
     In one embodiment of the invention, a communications system for transmitting and receiving signals comprises a baseband controller for coding a transmit signal and decoding a receive signal, a switch network coupled to the baseband controller, and surface acoustic wave (SAW) devices coupled to the switching network. On the transmitter side, a reference oscillator provides a plurality of reference signal pulses to the switch network. The SAW devices perform expander or correlator functions based on types of signals inputted. 
     In another embodiment of the invention, a surface acoustic wave (SAW) expander based transmitter comprises a baseband controller for coding an input signal and providing an output signal to a switching network, a reference oscillator for providing a plurality of reference signal pulses to the switching network, first and second chirp expanders coupled to the switching network for forming first and second chirped signals, and a combiner circuit for combining the first and second chirped signals into a transmit signal. 
     In another embodiment of the invention, a surface acoustic wave (SAW) correlator based receiver comprises: a circuit for receiving a signal, a splitter for splitting the received signal into first and second signals, first and second chirp correlators for compressing the first and second signals and forming first and second correlated pulses, and a baseband controller for decoding the correlated first and second pulses and forming a receive signal. 
     In yet another embodiment of the invention, a method of transmitting and receiving signals in a communications system comprises providing control signals to a switching network, and operating surface acoustic wave (SAW) devices coupled to the switching network. At a transmitter side, the method comprises generating a plurality of reference signal pulses and providing the pulses to the switching network. The SAW devices are operated at a radio frequency (RF) front end of the communications system. Active compensation schemes are not used. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a conventional surface acoustic wave (SAW) expander based transmitter. 
         FIG. 2  is a conventional SAW correlator based receiver. 
         FIG. 3  is an improved SAW expander based transmitter. 
         FIG. 4  is an improved SAW correlator based receiver. 
         FIG. 5  is a block diagram of chirp SAW correlator/expander functions necessary for a radio implementation. 
         FIG. 6  is a simplified schematic of SAW expander and correlator showing input and output signal transforms. 
         FIG. 7  is a diagram of slanted IDT. 
         FIG. 8  is a SAW based radio schematic transmitting up and down chirps. 
         FIG. 9  is a simplified SAW based radio schematic transmitting unidirectional chirps. 
         FIG. 10  is a conceptual layout of SAW correlator/expander (fundamental dimensions). 
     
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
     An embodiment of an improved surface acoustic wave (SAW) expander based transmitter  30  is shown in  FIG. 3 . The expander based transmitter  30  comprises a baseband section  31 , an oscillator  32 , two switches  33 , two chirp expanders  34 ,  35  (of opposite chirp direction) and a combiner  36 . Data is suitably coded in the baseband section  31  and its output controls the switches  33 . Data is imparted to the transmitter  30  by creating a train of pulses from the reference oscillator  32 . The pulses are fed to the chirp expanders  34 ,  35  to create an up or down ramping frequency pulse. The expanders  34 ,  35  are fabricated on a surface acoustic wave (SAW) substrate with novel interdigital transducers (IDTs). One IDT is used to implement programmable coding for multiple-access and the other is used to implement the chirp. After passing through the expanders  34 ,  35 , the chirped signals are then combined in combiner  36  to an antenna  37  for transmission. 
     Similarly, an improved SAW correlator-based receiver  40  is shown in  FIG. 4 . The receiver  40  comprises a splitter  46 , two chirp correlators  44 ,  45  (of opposite chirp direction), adaptive starting gating blocks  43 , demodulation circuits  42 , and a baseband section  41 . Following the antenna  47 , the received signal is split into two signals. The chirp correlators  44 ,  45  will then compress the signals according to the characteristics of the chirp design. Signals matching the direction and characteristics of the chirp correlators  44 ,  45  will result in a correlated output. All other signals will be uncorrelated. The chirp correlators  44 ,  45  are implemented using a SAW device with IDTs designed for coding and chirp functions, similar to the SAW expanders  34 ,  35 . The chirped transducer in the correlator is broken into several sub-bands to allow adaptive interference mitigation. This is simply implemented by breaking the IDT bus bar into sections. The sub-bands (sections) can be combined or deleted to mitigate interference or jamming. This function is depicted by the adaptive gating block  43  in  FIG. 4 . Following this block, the signals can be demodulated in demodulator  42  using a simple threshold detector. The detected output can then be decoded by the baseband section  41 . 
     Together,  FIGS. 3 and 4  show an improved system implementation of a chirp SAW correlator and expander in a dual channel receiver and transmitter. In the improved system implementation, the SAW devices are located directly at the RF front end instead of at an intermediate frequency (IF). In addition, no local oscillator (LO) or up/down converter is required. Furthermore, the SAW expander based receiver  40  requires no reference oscillator at all. 
       FIG. 5  shows a simplified block diagram of the chirped SAW correlator/expander functions necessary to implement the transmitter  30  and the receiver  40  in  FIGS. 3 and 4 . The transmitter  30  comprises two expanders  34 ,  35 , one with up-chirp (increasing frequency) and one with down-chirp (decreasing frequency), representing a logical 1 or 0, respectively. In an alternate embodiment, the up-chirp may represent logic ‘0’ and the down-chirp may represent logic ‘1’. Similarly, the receiver  40  comprises two correlators  44 ,  45  performing the correlation function. The receiver SAW devices are paired with those in the transmitter, such that the receiver has the inverse time response of the transmitter. 
     Advantages of the improved architecture include the ability of chirp to compensate for temperature drift and process variances by using chirp to sweep over a band of frequencies sufficient to overcome the expected temperature drift and manufacturing tolerances of the SAW device. This eliminates the need for active compensation schemes such as down conversion and tracking LOs used in conventional non-dispersive architectures. Consequently, the correlator/expander can be implemented directly at the RF front end, resulting in very low power consumption. In addition, chirp improves the impedance match of the devices, thereby improving insertion loss. 
     Another unique advantage of the improved architecture includes the ability to combine phase coding and chirp IDTs. A SAW correlator/expander comprises an input and output IDT. The combination of one coded transducer and one chirped transducer to achieve a coded chirped signal in a single device has the advantage of size reduction and lower insertion loss compared with two devices implementing these functions separately. The finger length of the proposed IDTs can be adjusted using a suitable apodization to modify the envelope characteristic for maximum sidelobe suppression. Apodization is a well-known window function applied in communication theory. In the case of a SAW device, the window function can be implemented by adjusting the finger length across the IDT according to a suitable window (apodization) function. 
       FIG. 6  shows two SAW devices  50 ,  60  and depicts the physical implementation of the correlators  44 ,  45  of receiver  40  and expanders  34 ,  35  of transmitter  30 . The input IDTs  51 ,  61  are phase coded with N number of symbols to allow multiple access coding (CDMA). The output IDTs  52 ,  62  are chirped in symmetrically opposite directions and broken into M sub-bands, which can be summed or deleted for interference/jamming mitigation. The two SAW devices  50 ,  60  depicted in  FIG. 6  are capable of performing all four of the correlator and expander functions outlined in  FIG. 5 . The function of each SAW device depends on the type of signal that is presented to its input terminals (on the left). If a pulse is presented to the input terminals of either SAW device  50 ,  60 , the device will perform as an expander and a chirped signal will appear at the output terminals. In this case, the SAW devices  50  and  60  will have up and down chirps, respectively. Similarly, if upchirp and downchirp signals are presented to the inputs of the SAW devices  50  and  60 , respectively, the outputs will be correlated pulses. That is, the device performs the function of a correlator. Any other signal combinations will result in uncorrelated outputs. 
       FIG. 7  shows a further unique feature of the improved architecture, which uses slanted fingers  71  in the phase coded IDT  70  to improve bandwidth. The 4 dB bandwidth of an IDT is approximately 1/Np (where No is the number of finger pairs). Therefore, for phase coded IDTs with constant finger spacing, the longer the code, the narrower the bandwidth of the IDT. This dependency can be overcome by using slanted fingers. Slanted IDTs have been demonstrated with maximum slant angles of up to 7 degrees for standard (non-coded) IDTs, but have not been applied to coded IDTs. The coding of the IDT  70  is achieved by the periodicity of finger connections to the bus bars  72 ,  73 . The upper  72  and lower  73  bus bars of each symbol (group of fingers) are exactly out of phase, so that driving the bus bars with the appropriate polarity will create a phase inversion in the signal representing coded symbols. 
       FIG. 8  shows a preferred embodiment of a complete transmitter/receiver radio architecture  80 . In particular, the architecture shows the unique use of the SAW correlator/expander  81  for both transmit and receive functions, depending on the signal that is switched to the input of the device  80 . Note that F and F in  FIG. 8  represent differential signal inputs. The switch network  82  includes the switching necessary to implement the programmable symbol coding and adaptive interference mitigations functions by controlling the signal connections to the IDT bus bars. 
       FIG. 9  shows an alternate simplified embodiment of the improved architecture. The radio architecture of  FIG. 9  employs unidirectional chirp only, to simplify the switching networks. Either up or down chirp may be implemented. The architecture uses each SAW device for either transmit or receive, but not both (unlike  FIG. 8 ). Thus, the overall data throughput for  FIG. 8  will be halved, all other things being equal, and an alternate signaling methodology would be required since the implementation of  FIG. 9  has only downchirps (or upchirps). Any standard signal protocol (for example, on-off keying (OOK)) may be used to signify a ‘1’ or ‘0’.  FIG. 9  may implement bi-direction chirps by duplicating the entire architecture with opposite chirp. A common baseband controller  95  may be used. 
     A further enhancement of the improved architecture is operation of the SAW devices at a harmonic frequency. Harmonic operation has the advantage of reducing lithographic demands, especially for high operating frequencies (for example, those greater than 800 MHz). In a preferred embodiment, the improved architecture operates at the third harmonic, but the use of other harmonics may also be practical. Harmonic operation also improves the impedance match of the devices (in addition to that from chirp), thereby improving insertion loss. Harmonic operation for SAW correlators/expanders in radio transmitter and receiver applications reduces lithographic demands and improves insertion loss and is unique. 
       FIG. 10  is a conceptual layout of a preferred embodiment of the SAW expander/correlator  100  along with the preferred dimensions.  FIG. 10  provides a more detailed representation of the SAW device  50  in  FIG. 6 . The layout and dimensions (and related discussion) of the embodiment are referenced to a fundamental center frequency of 2441.75 MHz. This can be easily transformed into a third harmonic design as the final step by simply omitting every second and third split electrode pair and tripling the width/spacing of the remaining fingers. 
     The design of the SAW device  60  in  FIG. 6  is very similar to the expander/correlator  100  shown in  FIG. 10 , except with opposite chirp. All other aspects of the device are identical and, therefore, not repeated here. The substrate is assumed to be X112° rotY-LiTaO 3 , with v=3276.6 m/s, though other piezoelectric substrates may be used in alternate embodiments. 
     Referring back to  FIG. 10 , the input IDT  110  is split into 32 symbols to allow phase coding. In the embodiment shown in  FIG. 10 , a 32 symbol code is used. In another embodiment, N symbol code may be used, where N is an integer. The bus bar  112  is split between symbols  120  to permit the polarity to be arbitrarily set (programmed) with an external switching network. Each symbol  120  is chosen to be 24λ long. Ideally, a smaller symbol length would be preferred since the design is severely pad limited. However, 24λ is 32.2 microns long, which is approximately the smallest practical staggered pad pitch supported by modern wire bonding equipment. The symbol size must be exactly divisible by 3λ to allow scaling to the third harmonic as the final step, if desired. 
     The input IDT  110  uses a split finger design (4 fingers per λ) to minimize reflections. The line/spacing dimension (Wc) is λ/8 or 0.1677 microns at the center frequency (2441.75 MHz), which is beyond current high volume manufacturing limits. As mentioned above, this may be mitigated in the final layout by omitting every second and third tap and tripling the width/spacing of the remaining taps to produce a third harmonic design. 
     The center frequency wavelength is given by λ=v/f=1.342 microns, where v=3276.6 m/s for X112° rotY-LiTaO 3  and f=2441.75 MHz. Therefore, the nominal length of the input IDT  110 =32 symbols×24λ long=1.0306 mm. 
     Unfortunately, a regularly spaced input IDT cannot be used. The large number of taps limits the bandwidth of the IDT to approximately 1/Np (where Np is the number of finger pairs; each finger is implemented as a split finger in this case). This is the same as the total number of wavelengths and gives an approximate bandwidth of 1/768 or about 0.13%, much too narrow to pass 60 MHz (or 2.5%) desired for this embodiment. 
     A method to increase the inherent bandwidth of an IDT is to use slanted fingers, as shown in  FIG. 7 . The fingers  71  are slanted to a common focal point, so that there is a constant minimum width (Wmin) at the top of the IDT fingers and a constant maximum width (Wmax) at the bottom end. These two widths are then set to the upper and lower frequencies, respectively. In this way, the fingers  71  are resonant at some point along their length across the desired bandwidth. 
     The width/spacing of the slanted IDT is given by the equations:
 
 W max=(min)/8 =v/ 8 f (max)
 
and
 
 W min=λ(max)/8 =v/ 8 f (min),
 
such that the minimum width/spacing Wmin=0.1657 microns and the maximum width/spacing Wmax=0.1698 microns at the fundamental frequency in  FIG. 7 .
 
     The slanted IDT  70  in  FIG. 7  trades the effective aperture for bandwidth since the IDT is resonant only over a small part of the IDT finger  71 . This effectively increases insertion loss. However, the reduced capacitive loading should improve the impedance match of the device to somewhat offset this. 
     Referring back to  FIG. 10 , the output IDTs  130  of the two SAW devices used for this system are a mirror image of each other, one with upchirp and one with downchirp. The finger spacing, Wn, on the output increases or decreases linearly, depending on the direction of chirp. The bus bars  132  on the output transducers  130  for this embodiment are broken into 16 sub-bands (equivalent to 3.75 MHz) to facilitate adaptive interference cancellation. The sub-bands  140  can be summed or deleted in response to the jamming environment. As noted earlier, the four correlator/expander functions can be accomplished using just two SAW devices because the SAW correlators used for the receiver can also be used to transmit (expand) the opposite channel in the transmitter. However, in the case of transmission, the sub-bands of the chirped IDT are all combined into one in the embodiments of  FIGS. 3 ,  8 , and  9 . In an alternate embodiment, the sub-bands may be summed or deleted in order to avoid transmitting in sub-bands known to include interference or jammers. 
     The length of the dispersive IDT sets the length of the frequency ramp (chirp length), whereas the delay is set by the distance between the centers of the input and output IDTs. For convenience, we choose the length of the input and output IDTs to be the same (i.e., 768λ). This number of wavelengths is convenient for the output IDT  130 , since the number is readily divisible into 16 sub-bands and also scales with an integer relationship to the third harmonic. It can be shown that the total length of the chirp IDT is the same as for a non-chirp input IDT (1.0306 mm), corresponding to a chirp time of 0.3145 microseconds. 
     The data rate is given by the reciprocal of the delay. For a 3 Mb/s data rate, a delay of ⅓ microsecond is required, which approximately corresponds to a distance:
 
( v*t )=1.0922 mm,
 
while the total delay path length is given by:
 
Total delay path length=½(Input  IDT  length+Output  IDT  length)+ IDT  Gap length.
 
Therefore, for a ⅓ microsecond delay and equal IDT lengths (1.0306 mm), the IDT gap is: 1.0922−½(1.0306+1.0306)=0.0616 mm.
 
     The selection of apodization and IDT aperture size is adjustable depending on the desired specifications. In this embodiment, the combined IDT response has a response equivalent to a Hamming function with 42 dB minimum sidelobe suppression. 
     Due to the high operating frequency, care must be taken with the physical layout to minimize RF losses, parasitics, and undesired coupling. The output traces and bond pads should be kept as short as possible to minimize the series inductance. Mutual coupling between adjacent I/Os should be minimized. In general, good RF practice should be exercised in the overall layout methodology to minimize the parasitic effects of the I/O. Special attention must be paid to parasitic coupling between bond pads and to traces passing between bond pads. 
     It should be appreciated that reference throughout this specification to “one embodiment” or “an embodiment” means that a particular feature, structure or characteristic described in connection with the embodiment is included in at least one embodiment of the present invention. Therefore, it is emphasized and should be appreciated that two or more references to “an embodiment” or “one embodiment” or “an alternative embodiment” in various portions of this specification are not necessarily all referring to the same embodiment. Furthermore, the particular features, structures or characteristics may be combined as suitable in one or more embodiments of the invention. 
     Similarly, it should be appreciated that in the foregoing description of exemplary embodiments of the invention, various features of the invention are sometimes grouped together in a single embodiment, figure, or description thereof for the purpose of streamlining the disclosure aiding in the understanding of one or more of the various inventive aspects. This method of disclosure, however, is not to be interpreted as reflecting an intention that the claimed invention requires more features than are expressly recited in each claim. Rather, as the following claims reflect, inventive aspects lie in less than all features of a single foregoing disclosed embodiment. Thus, the claims following the detailed description are hereby expressly incorporated into this detailed description, with each claim standing on its own as a separate embodiment of this invention.