Abstract:
An LLC series resonant converter and a driving method for the synchronous rectification power switches thereof are provided. The LLC series resonant converter includes a bridge circuit, a resonant network, a transformer, a rectification circuit, and a frequency adjustment controller. The bridge circuit includes at least one pair of power switches. The power switches drive the resonant network. The rectification circuit includes at least one pair of synchronous rectification power switches. The synchronous rectification power switches and the power switches have a mapping relation between them. The frequency adjustment controller provides driving signals to the synchronous rectification power switches in response to the operating frequency of the LLC series resonant converter and the series resonant frequency of the resonant network, to implement synchronous rectification in the LLC series resonant converter.

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
   This application claims the priority benefit of Taiwan application serial no. 94104986, filed on Feb. 21, 2005. All disclosure of the Taiwan application is incorporated herein by reference. 
   BACKGROUND OF THE INVENTION 
   1. Field of Invention 
   The present invention relates to an LLC series resonant converter and the driving method for the synchronous rectification power switches, and particularly to an LLC series resonant converter adopting a frequency adjustment controller to provide driving signals and the driving method of the synchronous rectification power switches thereof. 
   2. Description of Related Art 
     FIG. 1  is a prior art circuit configuration of a widely used LLC series resonant converter (SRC) having a half-bridge structure. Here, LLC represents a circuit composed by two inductors and a capacitor. It includes a half-bridge circuit ( 110 ) formed by a pair of power switches S 1  and S 2  for driving a resonant network ( 120 ). The resonant converter has 3 resonant parameters including a series resonant inductor Ls, a series resonant capacitor Cs and a magnetizing inductor Lm of a transformer TX (not shown in  FIG. 1 ). The series resonant inductor Ls can also be formed by leakage inductance from the transformer TX. Such three resonant parameters determine two characteristic resonant frequencies fs and fm shown in following: 
   
     
       
         
           
             
               
                 fs 
                 = 
                 
                   1 
                   
                     2 
                     ⁢ 
                     π 
                     ⁢ 
                     
                       
                         
                           L 
                           s 
                         
                         · 
                         
                           C 
                           s 
                         
                       
                     
                   
                 
               
             
             
               
                 ( 
                 1 
                 ) 
               
             
           
           
             
               
                 fm 
                 = 
                 
                   1 
                   
                     2 
                     ⁢ 
                     π 
                     ⁢ 
                     
                       
                         
                           ( 
                           
                             
                               L 
                               s 
                             
                             + 
                             
                               L 
                               m 
                             
                           
                           ) 
                         
                         · 
                         
                           C 
                           s 
                         
                       
                     
                   
                 
               
             
             
               
                 ( 
                 2 
                 ) 
               
             
           
         
       
     
   
   The input terminal of the LLC series resonant converter is a DC voltage Vin. The transformer TX isolates the half-bridge circuit ( 110 ) and the resonant network ( 120 ) from a rectification circuit ( 130 ) by a primary side winding n p  and two series secondary sides winding n s1  and n s2 , coupled in series. 
   The rectification circuit ( 130 ) includes a pair of rectification diodes D 1  and D 2  connected to an output capacitor Co. The cathodes of D 1  and D 2  are connected to the anode of capacitor Co. The anode of D 1  is connected to the positive terminal of the secondary side coil ns 1 , whereas the anode of D 2  is connected to the negative terminal of the secondary side coil ns 2 . A common connection node between the coils ns 1  and ns 2  is a ground of an output voltage Vo. 
   The power switches S 1  and S 2  of the LLC series resonant converter operate under the condition of equal pulse width, that is, in the condition of 50%. A frequency adjustment controller  140  is therefore demanded because the adjustment of the output voltage is obtained by changing the operation frequency. 
   LLC series resonant converter has the characteristics that proper parametric design and operating ranges can guarantee the power switches of a primary side bridge circuit to be operated under a zero voltage switching (ZVS) condition, and at the same time causing the rectification switches of a secondary side to perform as zero current switching (ZCS). 
     FIG. 2  is a schematic diagram of the operating situation time sequence of the LLC series resonant converter shown in  FIG. 1 , when the operating frequency f of power switches S 1  and S 2  satisfies the formula of
   fm≦f≦fs   (3), 
wherein V g,S1  and V g,S2  respectively refer to the driving signals for the power switches S 1  and S 2 .
 
   At time of t 0 , because the primary side current i r  is opposite to a reference direction, the power switch S 1  opens under the ZVS condition. During the interval between t 0  and t 1 , the rectification diode D 1  is conducted with current, therefore the voltage on the magnetizing inductor Lm is constant and does not take part in the resonance, so that the magnetizing current i m  increases linearly. Referring to  FIG. 2 , due to the resonance between Ls and Cs, current i D1  through D 1  appears in a quasi-sine waveform. 
   At time of t 1 , because the switching period is longer than the resonant period between Ls and Cs, i r  descends down to i m  before the shutting off of S 1 , so D 1  is off at this moment. Cs, Ls and Lm take part in the resonance. For the purpose of simplifying analysis, supposed to be the condition of Lm&gt;&gt;Ls, then the i r  is approximately a straight line. 
   At time of t 2 , S 1  is turned off and S 2  is conducted. At time of t 3 , S 2  is switched on under ZVS condition. 
   During the intervals from t 3  to t 4  and from t 4  to t 5 , similar process can be analyzed out. The operation status and current waveform i D2  similar to D 1  also happen to the rectification diode D 2 . i D1  and i D2  compose the output rectification current i rec . 
   If the LLC series resonant converter operates at switch frequency fs, the dead time of output rectification current i rec , in which both rectification diodes D 1  and D 2  are non-conducting during the dead time, disappears and rectification current i rec  has a quasi-sine absolute value waveform. 
   In order to operate in a wide range of input voltage and output load, the LLC series resonant converter in practical operation is usually operating under the condition that the dead time the i rec  is zero. As, at this time, the shutting off action of the rectification diode D 1  or D 2  always happens before the power switches S 1  or S 2  are off, the conducting pulse widths of D 1  and D 2  are smaller than those of S 1  and S 2 . Nowadays, it becomes popular to adopt a power metal oxide semiconductor field effect transistor (MOSFET) to substitute rectification diodes D 1  and D 2  operating as synchronous rectification power switches. In this manner, the driving pulse of the synchronous rectification power switches has to be off when the backward current flowing from the source to the drain descends down to zero, that is, non-conducting state during the dead time of the i rec . Otherwise, the output terminal of the converter would transport power to the primary side of the converter, and the circuit would not be able to operate properly and safely. Accordingly, the driving signals of power switches S 1  and S 2  cannot be used to obtain the driving signals for synchronous rectifiers. 
   Moreover, the coil of the transformer TX could be used to obtain the driving signals for the synchronous rectifier power switches, since during the dead time of i rec , the voltage of secondary side coil is not zero, but is the resonant voltage on Lm. 
   Therefore, better technique is demanded to solve the above problems to provide proper driving signals, so that the power MOSFET can replace the rectification diodes and the synchronous rectification in the LLC series resonant converter can be implemented, and it can guarantees the safe and reliable operation under any operation condition. 
   SUMMARY OF THE INVENTION 
   It is an object of the present invention to provide an LLC series resonant converter, and a driving method for the synchronous rectification power switches to obtain suitable driving signals, which allow the power MOSFETs to substitute the rectification diodes, therefore synchronous rectification in the LLC series resonant converter can be implemented, and operation under any conditions can be guaranteed in safe and reliability. 
   For achieving the foregoing object and others, the present invention provides an LLC series resonant converter, which includes a bridge circuit, coupling to an input voltage, and including at least one pair of power switches; a resonant network, coupling to the bridge circuit, being driven by the power switches; a transformer, coupling to the resonant network; a rectification circuit, coupling to the transformer, the transformer providing output current of the LLC series resonant converter. The rectification circuit includes at least one pair of synchronous rectification switches. The synchronous rectification power switches and the power switches have a mapping relation between them. The synchronous rectification switch and its corresponding power switch are synchronously on or off. A frequency adjustment controller is coupled between the power switches and the synchronous rectification power switches, supplying driving signals for the synchronous rectification power switches, and thereby implementing synchronous rectification in the LLC series resonant converter. When the operating frequency of the LLC series resonant converter is less than the series resonant frequency of the resonant network, the driving signals of each of the synchronous rectification switches are constant width pulse signals which are synchronous with their corresponding power switch&#39;s signals. The width of the constant width pulse signals is determined by the series resonant parameter of the resonant network. When the operating frequency of the present invented LLC series resonant converter is higher than or equal to the series resonant frequency of the resonant network, the driving signals of each of synchronous rectification power switches are the driving signals of the corresponding power switches. 
   From another aspect, the present invention is to provide a driving method for the synchronous rectification power switches, which is used in an LLC series resonant converter. The LLC series resonant converter includes a bridge circuit, a resonant network, and a rectification circuit, wherein the bridge circuit includes at least one pair of power switches. The rectification circuit includes at least one pair of synchronous rectification power switches. One of synchronous rectification power switches Q 1  and Q 2  and one of the power switches S 1  and S 2  are correspondingly and electrically coupled through one of a pair of driving signal generators. The corresponding synchronous rectification power switches and the power switches are synchronously on or off. The driving method for the synchronous rectification power switches includes the steps of: obtaining the operating frequency of the LLC series resonant converter; obtaining the series resonant frequency of the resonant network; and when the operating frequency of the LLC series resonant converter is lower than the series resonant frequency of the resonant network, the driving signals of each of the synchronous rectification switches being constant width pulse signals which are synchronous with their corresponding power switch&#39;s signals. The width of the constant width pulse signals is determined by the series resonant parameter of the resonant network. Also and, when the operating frequency of the LLC series resonant converter is higher than or equal to the series resonant frequency of the resonant network, the driving signals of each of the synchronous rectification power switches are the driving signals of the corresponding power switches. 
   The present invention is based on the relation in quantity between the operating frequency of the LLC series resonant converter and the series resonant frequency of the resonant network, and based on the diving signals of the power switches, so as to provide the driving signals of the synchronous rectification power switches. As a result, the power MOSFET can replace the rectification diodes and implement the synchronous rectification in the LLC series resonant converter, and further assure the safe and reliable operation in any operation condition. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The accompanying drawings are included to provide a further understanding of the invention, and are incorporated in and constitute a part of this specification. The drawings illustrate embodiments of the invention and, together with the description, serve to explain the principles of the invention. 
       FIG. 1  is a conventional circuit configuration of a widely used LLC series resonant converter. 
       FIG. 2  is a schematic diagram of the operating situation sequence of the conventional LLC series resonant converter. 
       FIG. 3  is a circuit configuration of an LLC series resonant converter, according to a preferred embodiment of the present invention. 
       FIG. 4  and  FIG. 5  are timing sequence, schematically illustrating the timing of the main signals in operating the LLC series resonant converter, according to a preferred embodiment of the present invention. 
       FIG. 6  is a flowchart, schematically illustrating the driving method for the synchronous rectification power switches, according to a preferred embodiment of the present invention. 
       FIG. 7  is a circuit configuration, schematically illustrating another LLC series resonant converter, according to another preferred embodiment of the present invention. 
   

   DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   A following preferred embodiment is taken to describe the LLC series resonant converter (SRC). 
   According to the operating waveform of the LLC series resonant converter, as shown in  FIG. 2 , when the output rectification current i rec  has a dead time, the conducting time of the rectification diodes D 1  and D 2  is a half time period 1/(2fs) of the LLC series resonant converter, but it is determined by the values of Ls and Cs. When a range of the input voltage or the output voltage changes, provided fm&lt;f&lt;fs, the conducting time of each of the rectification diodes D 1  and D 2  does not change thereafter. The change is just the dead time of the rectification current i rec , which is the time interval from t 1  to t 2  or from t 4  to t 5 . When the operating frequency is higher than fs, the output rectification current i rec  does not have a dead time. Therefore, the conducting times of D 1  and D 2  are respectively corresponding to the conducting times of the power switches S 1  and S 2 . 
   In view of the above, the preferred embodiment adopts a self-adaptive synchronous rectification driving method for the LLC series resonant converter, wherein the pulse width of the driving signals can be automatically adjusted according to the operating frequency of the LLC series resonant converter. When the operating frequency is between fm and fs, the synchronous rectification driving signals have a constant pulse width. When the frequency of the LLC series resonant converter is adjusted due to the change of the input voltage or the output load, the synchronous rectification driving signals therefore synchronously change their frequencies, while the pulse width does not change. When the operating frequency is higher or equal to fs, the synchronous rectification driving signals is determined by the driving signals of the power switches S 1  and S 2 , being in-phase and having the same pulse width. Consequently, the pulse width of the synchronous rectification switches changes according to the operating frequency. 
     FIG. 3  schematically illuminates the operation mechanism of a preferred embodiment of the present invention for the LLC series resonant converter. The devices therein includes a half-bridge circuit  110  coupled to an input voltage V in , a resonant network  120  driven by the power switches S 1  and S 2 , a transformer TX, a rectification circuit  310  for providing current from the transformer TX, and a frequency adjustment controller  320 . The preferred embodiment of the present invention is different from the conventional circuit shown in  FIG. 1  by rectification circuit  310  for replacing the conventional rectification circuit  130 , wherein the synchronous rectification power switches Q 1  and Q 2  has replaced the conventional rectification diodes D 1  and D 2 . Also and, a preferred frequency adjustment controller  320  has replaced the conventional frequency adjustment controller  140 . The frequency adjustment controller  320  includes one pair of driving signal generators  330  and  340 , a locking circuit  250  and a voltage control oscillator (VCO)  350 . The driving signal generators  330  further includes a synchronous signal circuit  210 , a constant width pulse generator  220  and an AND logic gate  360 . The driving signals generator  340  further includes a synchronous signal circuit  230 , a constant width pulse generator  240  and an AND logic gate  370 . The relation between the devices will be described later. As can be seen in  FIG. 3 , one of synchronous rectification power switches Q 1  and Q 2  and one of the power switches S 1  and S 2  are correspondingly and electrically coupled through one of the pair of driving signal generators  330  and  340 . 
   For easy descriptions, a common source connecting manner for the synchronous rectification switches Q 1  and Q 2  is taken, but it is not the only choice in actual application. 
   As shown in  FIG. 3 , after the driving signals V g,S1  and V g,S2  of the power switches S 1  and S 2  are respectively processed by the synchronous signal circuits  210  and  230 , the synchronous signals synchronizing with the driving signals V g,S1  and V g,S2  are generated, and then the rising edges of the synchronous signals respectively trigger the constant width pulse generators  220  and  240 . As a result, the constant width pulse generators  220  and  240  export two constant width pulse signals V pulse1  and V pulse2  which are respectively synchronous to the rising edges of the signals V g,S1  and V g,S2 . The pulse width is preferably set as a half period 1/(2fs) or slightly smaller where fs is the series resonant frequency of the resonant network  120 . After the constant width pulses V pulse1  and V pulse2  have been respectively logically operated with the signals V g,S1  and V g,S2  by the AND logic gates  360  and  370 , the output signals of V g,Q1  and V g,Q2  can be used for providing the driving signals to the synchronous rectification power switches Q 1  and Q 2 . 
     FIG. 4  is a timing sequence diagram of synchronous rectification driving for the LLC series resonant converter, when the operating frequency is between fm and fs, according to the embodiment of the present invention. In this situation, since the width of V pulse1  and V pulse2  is smaller than that of V g,S1  and V g,S2 , after AND logic operation, the output is determined by V pulse1  and V pulse2  as two constant width pulse driving signals V g,Q1 and V   g,Q2 . 
     FIG. 5  is a timing sequence diagram of synchronous rectification driving for the LLC series resonant converter, when the operating frequency is higher than fs, according to the embodiment of the present invention. In this situation, since the width of V pulse1  and V pulse2  is larger than that of V g,S1  and V g,S2 , after a AND logic operation, the output results of V g,Q1  and V g,Q2  are consistent with V pulse1  and V pulse2 , and the pulse width changes, according to the changes of V g,S1  and V g,S2 . 
   For the purpose of preventing the synchronous rectification switches Q 1  and Q 2  from being difficultly driven in operating at an over-high frequency, usually, when the LLC series resonant converter soft starts or operates in unloaded status under no load condition, the locking circuit  250  would judge according to the operating frequency control signals V EA . Then a locking signal is issued to lock the output from the constant width pulse generators  220  and  240  V g,Q1  and V g,Q2 , which are thereby locked under the foregoing condition, so that the synchronous rectification switches Q 1  and Q 2  are at the off state. The operating frequency control signals V EA  can be the output of the voltage control oscillator  350  in the frequency adjustment controller  320 , or any other signals capable of indicating the operating frequency. 
   Although two AND logic gates  360  and  370  are used to implement the self-adaptive control of the synchronous rectification driving signals, the actual circuit implementation is not limited to this logic gate structure. Any circuits that substantially behave the similar function of AND logic gate are covered within the scope of the present invention. 
   According to  FIG. 6 , a second preferred embodiment is given to depict the driving method of the synchronous rectification power switches for the LLC series resonant converter of the present invention. The second preferred embodiment about the driving method is supposed to be used in the LLC series resonant converter of the above disclosed preferred embodiment. 
   First, a signal of the operating frequency of the LLC series resonant converter is obtained in step  602 , and a signal of series resonant frequency of the resonant network is obtained in step  604 . Then, in step  606 , it is judged whether or not the operating frequency of the LLC series resonant converter is excessive, causing difficulty in driving the synchronous rectification switches. If it is, then, in step  608 , a zero-voltage driving signal is provided to the synchronous rectification power switches to keep the synchronous rectification power switches in off state. This is to protect the synchronous rectification power switches from damage. 
   If the operating frequency is not excessive in step  606 , in the following step  610 , it is judged whether or not the operating frequency of the LLC series resonant converter is higher or equal to the series resonant frequency of the resonant network. If it is, in step  612 , the driving signals corresponding to the power switches are provided to serve as the driving signal for the synchronous rectification power switches. Otherwise, in step  614 , the constant width signals, which are synchronous with the driving signals of the corresponding power switches and have width determined by the resonant parameter of the resonant network, are provided to serve as the driving signals of the synchronous rectification power switches. 
     FIG. 7  depicts a third preferred embodiment of the present invention, showing a circuit configuration of the LLC series resonant converter. The main difference between  FIG. 7  and  FIG. 3  is substituting the half-bridge circuit  110  in  FIG. 3  with a full bridge circuit  710  in  FIG. 7 . The full bridge circuit  710  includes four power switches marked as S 1  to S 4 , wherein S 1  and S 4  have the same driving signals, and S 2  and S 3  have the same driving signals. The operation and connection relation of the other parts in  FIG. 7  are substantially the same as the corresponding part in  FIG. 3 . 
   In addition, although the rectification  310  is preferred in the above described preferred embodiment as shown in  FIG. 7 , the rectification circuits can also include more pair of the synchronous rectification power switches. The persons skilled in the related ordinary art can easily know the extension in the foregoing embodiment about the driving relation between the power switches for the half-bridge circuit or the full-bridge circuit, and the synchronous rectification power switches of the rectification circuit. 
   As illustrated above, the present invention is based on the quantities of the operating frequency for the LLC series resonant converter and the series resonant frequency for the resonant network as well as the driving signals for the power switches, so as to provide the driving signals for the synchronous rectification power switches. Therefore, the power MOSFET is allowed to replace the conventional rectification diodes, and the synchronous rectification in the LLC series resonant converter can be implemented, and the operation can be assured in safe and reliability at any operating condition. 
   Other modifications and adaptations of the above-described preferred embodiments of the present invention may be made to meet particular requirements. This disclosure is intended to exemplify the invention without limiting its scope. All modifications that incorporate the invention disclosed in the preferred embodiment are to be construed as coming within the scope of the appended claims or the range of equivalents to which the claims are entitled.