Abstract:
An improved phase noise tracker comprising a first rotator, delayed second rotator and feedback loop coupled to the first and second rotators. The feedback loop further comprises a phase error detector and low-pass filter. The phase error detector estimates a phase error value of the first rotator&#39;s output, and the low-pass filter smooths out the output of the phase error detector by accumulating previous estimated phase error values from the phase error detector. The output of the feedback loop, from the low-pass filter&#39;s output, is fedback to a phase control input of the first rotator to control the phase rotation of the first rotator. The feedback loop&#39;s output is fed to a phase control input of the delayed second rotator to control its phase rotation. Therefore, the improved phase noise tracker tracks phase noise based on both previous and future phase error values, which more accurately corrects for phase noise.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    This invention relates to the correction of phase noise in a system and, more particularly, to a phase noise tracker for correcting phase noise using a delayed rotator.  
           [0003]    2. Related Art  
           [0004]    A trellis-coded 8-VSB (Vestigial Sideband) signal format is a standard for terrestrial DTV (Digital TV) broadcasting that was approved by the ATSC (Advanced Television Systems Committee) in 1995. The 8-VSB signal format has 8 discrete data levels and is segmented into symbols, which are transmitted at a rate of about 10 mega-symbols per second.  
           [0005]    DTV receivers typically include several local oscillators. The local oscillators are used to generate sinusoidal signals to down-convert the frequency of an incoming DTV signal. Unfortunately, practical local oscillators do not produce a pure sinusoid but rather smeared sinusoidal signals that introduce phase noise into the DTV signal. Uncompensated phase noise can lead to long bursts of errors in a DTV trellis decoder, which substantially degrade the performance of the decoder. As a result, DTV receivers typically employ a phase noise tracker to correct for the phase noise.  
           [0006]    [0006]FIG. 1 shows a typical phase noise tracker  3  coupled to a Hilbert filter  7 . The Hilbert filter  7  enables the phase tracker  3  to track the phase noise of an incoming 8-VSB signal. The Hilbert filter  7  has an input  5  for receiving an incoming 8-VSB signal and a complex output  10   a  and  10   b , as understood by those of skill in the art and explained briefly below. The phase tracker  3  comprises a complex rotator  15  and a feedback loop  20 . The complex rotator  15  has a complex input coupled to the output  10   a  and  10   b  of the Hilbert filter  7 , a phase control input  17  and a complex output  19   a  and  19   b . The output of the phase tracker  3  is taken at the output  19   a  and  19   b  of the rotator  15 . The feedback loop  20  has a complex input coupled to the output  19   a  and  19   b  of the rotator  15  and an output  45  coupled to the phase control input  17  of the rotator  15 . The feedback loop  20  further comprises a phase error detector  30  and a low-pass filter  40 . The phase error detector  30  has a complex input coupled to the output  19   a  and  19   b  of the rotator  15  and an output  35 . The low-pass filter  40  has an input coupled to the output  35  of the phase error detector  30  and an output  45  coupled to the phase control input  17  of the rotator  15 .  
           [0007]    The Hilbert filter  7  is used to transform an incoming input 8-VSB signal at its input  5  into a complex signal having an I (in-phase) component  10   a  and a Q (quadrature) component  10   b . The complex signal  10   a  and  10   b  is sent to the rotator  15  of the phase tracker  3 . The rotator  15  rotates the phase of the complex signal  10   a  and  10   b  by an amount controlled by the output  45  of the feedback loop  20 , which is coupled to the phase control input  17 . The phase error detector  30  of the feedback loop  20  estimates the phase error of the output  19   a  and  19   b  of the rotator  15 . The phase error detector  30  then outputs an estimated phase error value on output line  35  to the low-pass filter  40 . The low-pass filter  40  smoothes out the output  35  of the phase error detector  30  by accumulating previous estimated error phase values from the error phase detector  30 . Thus, the output  45  of the low-pass filter  40  is based on previous estimated phase error values and slowly tracks changes in the estimated phase error value. The output  45  of the low-pass filter  40  is feed back to the rotator  15  at its phase control input  17 . This causes the rotator  15  to rotate the phase of the complex signal  10   a  and  10   b  in a direction that decreases the estimated phase error value, and thereby reduce phase noise.  
           [0008]    The phase error caused by phase noise is typically correlated to both previous and future phase error values. A drawback of the above phase tracker  3  is that the output  45  of the feedback loop  20 , which controls the phase rotation of the rotator  15 , is based on previous phase error values. Therefore, there is a need for a phase noise tracker that tracks phase noise based on both previous and future phase error values. This would allow the phase noise tracker to more accurately correct for phase noise and operate under more severe phase noise conditions.  
         SUMMARY  
         [0009]    This invention provides a phase noise tracker that corrects for phase noise and can operate under severe phase noise conditions.  
           [0010]    One embodiment of the improved phase noise tracker comprises the components of the phase tracker shown in FIG. 1 in addition to a complex delay element and a second complex rotator. The complex delay element is coupled between the Hilbert filter and the second complex rotator. The feedback loop circuit controls the phase rotation of the second rotator.  
           [0011]    The effect of the delay element is to delay the input of the second rotator with respect to the input of the feedback loop. This causes the estimated phase error value of the feedback loop to be ahead in time compared to the delayed input of the second rotator. As a result, the output of the feedback loop is based on both previous and future estimated phase error values relative to the second rotator. Because the output of the feedback loop is used to control the phase rotation of the second rotator, the second rotator is able to track phase noise based on both previous and future estimated phase error values.  
           [0012]    Therefore, the use of the delay and the second rotator enables the phase noise tracker to more accurately track phase noise based on both previous and future estimated phase error values. This allows the improved phase noise tracker to operate under more severe phase noise conditions than the prior art.  
           [0013]    Other systems, methods, features and advantages of the invention will be or will become apparent to one with skill in the art upon examination of the following figures and detailed description. It is intended that all such additional systems, methods, features and advantages be included within this description, be within the scope of the invention, and be protected by the accompanying claims. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWING  
       [0014]    The components in the figures are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention. Moreover, in the figures, like reference numerals designate corresponding parts throughout the different views.  
         [0015]    [0015]FIG. 1 illustrates a block diagram of a prior art phase tracker.  
         [0016]    [0016]FIG. 2 is a block diagram illustrating a phase tracker.  
         [0017]    [0017]FIG. 3 is a block diagram illustrating a Hilbert filter.  
         [0018]    [0018]FIG. 4 illustrates an I/Q diagram used in estimating phase error.  
         [0019]    [0019]FIG. 5 is a block diagram illustrating a low-pass filter.  
         [0020]    [0020]FIG. 6 is a block diagram illustrating an AGC (automatic gain control) feedback loop.  
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT  
       [0021]    [0021]FIG. 2 illustrates a phase tracker  200  according to an example embodiment of the invention. The phase tracker  200  comprises all the components of the phase tracker  3  shown in FIG. 1. In addition, the phase tracker  200  further comprises a complex delay  210  and a second rotator  220 . The complex delay  210  has a complex input coupled to the output  10   a  and  10   b  of the Hilbert filter  7  and a complex output  215   a  and  215   b . The second complex rotator  220  has a complex input coupled to the output  215   a  and  215   b  of the delay  210 , a phase control input  225  and a complex output  219   a  and  219   b . The output of the phase tracker  200  is taken at the output  219   a  and  219   b  of the second rotator  220 . The output  45  of the feedback loop  20  is coupled to the phase control input  17  of the first rotator  15  as well as the phase control input  225  of the second rotator  220 . Thus, the phase rotation of the first rotator  15  and the second rotator  220  are both controlled by the same feedback loop  20 .  
         [0022]    The complex signal  19   a  and  19   b  inputted to the feedback loop  20  is ahead in time compared to the delayed input signal  215   a  and  215   b  of the second rotator  220 . This causes the estimated phase error value  35  of the phase error detector  30  to also be ahead in time compared to the delayed input signal  215   a  and  215   b  of the second rotator  220 . As a result, the output  45  of the feedback loop  20  is based on both previous and future estimated phase error values relative to the second rotator  220 .  
         [0023]    Because the output  45  of the feedback loop  20  is used to control the phase rotation of the second rotator  220 , the second rotator  220  is able to track the phase noise of its delayed input signal  215   a  and  215   b  based on both previous and future estimated phase error values. As a result, the improved phase tracker  200  more accurately tracks phase noise than the phase noise tracker of the prior art. This allows the improved phase tracker  200  to operate in more severe noise conditions. In addition, for a given amount of phase noise, the improved phase tracking leads to a lower symbol error rate (SER) and hence better performance.  
         [0024]    The number of future estimated phase values relative to the delayed input signal  215   a  and  215   b  of the second rotator  220  is a function of the amount of delay introduced by the delay  210 . The greater the delay, the greater the number of estimated phase values. The amount of delay introduced by the delay element  210  is typically given in units of taps, which correspond to one symbol of the complex signal  10   a  and  10   b . Preferably, the delay element  210  introduces a delay of about 50 taps.  
         [0025]    The Hilbert filter  7 , the phase error detector  30  and the low-pass filter  40  according to an example embodiment of the invention will now be described in detail with reference to FIGS. 3, 4 and  5 . FIG. 3 illustrates a Hilbert filter  7  built in accordance with an example embodiment of the invention. The Hilbert filter  7  comprises fourteen 1-tap delays  310   a - 310   n  coupled in series and eight branches  320   a - 320   h , where each branch  320   a - 320   h  is taken between every other 1-tap delay  310   a - 310   n . For example, the first branch  320   a  is taken at the input  5  of the first 1-tap delay  310   a , the second branch  320   b  is taken between 1-tap delays  310   b  and  310   c , and the third branch  320   c  is taken between 1-tap delays  310   d  and  310   e . The Hilbert filter  7  also comprises eight multipliers  330   a - 330   h , wherein each multiplier  330   a - 330   h  multiplies one of the branches  320   a - 320   h  by a coefficient. The number next to each multiplier  330   a - 330   h  in FIG. 3 indicates an example value of the coefficient of the multiplier  330   a - 330   h . The Hilbert filter  7  further comprises an adder  340  for adding the outputs of the eight multipliers  330   a - 330   h.    
         [0026]    The input 8-VSB signal  5  is inputted to the first 1-tap delay  310   a . In the example embodiment, the I component  10   a  of the Hilbert filter  7  output is taken at the output of the seventh 1-tap delay  310   g . Therefore, the I component  10   a  is simply the input 8-VSB signal  5  delayed by 7 taps. The Q component  10   b  of the Hilbert filter  7  output in the example embodiment is taken at the output of the adder  340 , which approximates a Hilbert transform.  
         [0027]    The Hilbert filter  7  is used to generate a “virtual” 2-dimensional complex signal having an I component  10   a  and a Q component  10   b  from the input 8-VSB signal  5 . No new information is added by generating the “virtual” 2-dimensional complex signal. However, the “virtual” complex signal allows the phase of the input 8-VSB signal  5  to be estimated in a similar manner as a normal complex signal such as a QAM (Quad-Amplitude Modulated) signal.  
         [0028]    In FIG. 4, the phase error detector  30  is illustrated with an I/Q diagram  405  in which the I component is represented by a horizontal axis  410  and the Q component is represented by a vertical axis  420 . Eight vertical data lines  425   a - 425   g  intersect the horizontal axis  410 . Each data line  425   a - 425   g  intersects the horizontal line  410  at one of eight I component data values. For example, these eight data values may be −7, −5. −3, −1, +1, +3, +5 and +7, as shown in FIG. 4.  
         [0029]    Point A on the I/Q diagram  405  represents one symbol of the complex signal  19   a  and  19   b  inputted to the phase error detector  30 . The vertical position of point A represents the Q component  19   a  and the horizontal position of point A represents the I component  19   b . A line  430  extending from the origin  407  of the I/Q diagram  405  to point A provides the magnitude and the phase angle of point A. The length of line  430  gives the magnitude of point A and the angle between line  430  and the horizontal line  410  gives the phase angle of point A.  
         [0030]    To estimates the phase error of the complex signal  19   a  and  19   b  at point A, the phase error detector  20  first determines which one of the eight data values on the horizontal line  410  is closest to the I component  19   b  of point A. In this example case, the closest data value is +5. The phase error detector  30  then determines an angle VI needed to rotate line  430  about the origin  407  such that the end of line  430  touches or intersects the data line  425   f  of data value +5. Line  440  and point B represent line  430  and point A rotated clockwise by angle VI. Angle VI gives the estimated phase error value (on output line  35 ) of the complex signal  19   a  and  19   b  at point A. The phase error detector  30  follows a similar procedure to estimate the phase error value for each symbol of the complex signal  19   a  and  19   b.    
         [0031]    [0031]FIG. 4 also shows a line  450  extending from the data value +5 on the horizontal line  410  (e.g., the intersection of horizontal line  410  and data line  425   f ) to point A. The angle V 2  between line  450  and data line  425   f  can be used to approximate angle V 1  for small values of V 1 . The advantage of using angle V 2  to approximate angle V 1  is that the value of V 2  is easier to calculate than V 1 .  
         [0032]    [0032]FIG. 5 illustrates a low-pass filter  40  used with an example embodiment of the invention. The low-pass filter  40  comprises an adder  510 , a multiplier  520 , and a delay  530 . The adder  510  has a first input coupled to the output  35  of the phase error detector, a second input  512 , and an output  515  coupled to the output  45  of the low-pass filter  40 . The multiplier  520  has an input coupled to the output  515  of the adder  510  and an output  525 . The delay element  530  has an input coupled to the output  525  of the multiplier  520  and an output coupled to the second input  512  of the adder  510 .  
         [0033]    The multiplier  520  and the delay element  530  form a feedback loop that feeds a signal proportional to the output  515  of the adder  510  back to the second input  512  of the adder  510 . As a result, the adder  510  accumulates previous estimated phase error values. This enables the adder  510  to smooth out the output  35  of the phase error detector  30  based on previous estimated phase error values.  
         [0034]    The multiplier  520  multiplies the output  512  of the adder  510  by a leaking factor  526  having a value less than 1, preferably 0.90. This is done to slowly leak off the accumulated phase error value of the low-pass filter  40 . The delay  530  is used to delay the output  525  of the multiplier  520  so that it matches the arrival of estimated phase error values  35  from the phase error detector  30 .  
         [0035]    [0035]FIG. 6 shows two variable gain amplifiers  660   a  and  660   b  and an AGC (automatic gain control) feedback loop  615  coupled to the phase noise tracker  200 . The two amplifiers  660   a  and  660   b  and the AGC feedback loop  615  may be used to automatically adjust the amplitude of the complex output signal  10   a  and  10   b  of the Hilbert filter  7 . The gain of both amplifiers  660   a  and  660   b  are controlled by a gain control input  665 . One of the amplifiers  660   a  has an input coupled to the I component  10   a  of the Hilbert filter  7  output, and the other amplifier  660   b  has an input coupled to the Q component  10   b  of the Hilbert filter  7  output. Each amplifier  660   a  and  660   b  has an output  610   a  and  610   b , respectively, coupled to one of the complex inputs of the phase noise tracker  200 . The AGC feedback loop has an input coupled to the I component  19   a  output of the first rotator  15  and an output  650  coupled to the gain control input  665  of the amplifiers  660   a  and  660   b . Thus, the output  650  of the AGC feedback loop  615  controls the gain of both amplifiers  660   a  and  660   b.    
         [0036]    The AGC feedback loop  615  further comprises an AGC error detector  620  and a low-pass filter  640 . The AGC error detector  620  has an input coupled to I component output  19   a  of the first rotator  15  and an output  630 . The low-pass filter  640  has an input coupled to the output  630  of the AGC error detector  620  and output  650  coupled to the gain control input  665  of the amplifiers  660   a  and  660   b.    
         [0037]    In one example embodiment, the AGC error detector  620  compares the amplitude of the I component output  19   a  of the first rotator  15  to eight allowable I component data values. These eight allowable data values may, for example, be −7, −5, −3, −1, +1, +3, +5 and +7. The AGC error detector  620  determines which one of the allowable data values is closest to the amplitude of the I component output  19   a  of the first rotator. The AGC error detector  620  then outputs the difference between the closest allowable data value and the amplitude of the I component output  19   a  of the first rotator  15  as an estimated AGC error  630 . The estimated AGC error  630  is inputted to the low-pass filter  640 , which smoothes out the estimated AGC error  630 . The lowpass filter  640  may be similar to the low-pass filter  40  used with the phase noise tracker  200 . The filtered estimated AGC error  650  is then inputted to the gain control input  665  of the amplifiers  660   a  and  660   b . This causes amplifier  660   a  to adjust the amplitude of the I component output  19   a  of the first rotator  15  in a direction that reduces the estimated AGC error  630 .  
         [0038]    While various embodiments of the application have been described, it will be apparent to those of ordinary skill in the art that many more embodiments and implementations are possible that are within the scope of the subject invention.  
         [0039]    For example, even though the invention has been described in the context of a DTV receiver, those skilled in the art will appreciate that the invention can be implemented in a variety of systems requiring phase noise correction, especially for phase noise created by a local oscillator. Such systems include, but are not limited to, cable modems and GPS (Global Positioning Systems) receivers. In addition, those skilled in the art will appreciate that the invention can be implemented in systems using QAM (Quad-Amplitude Modulated) signals and PSK (Phase Shift Key) signals, as well as 8-VSB signals. For systems using QAM signals and PSK signals, the Hilbert filter can be omitted, since these signals are already complex signals. Therefore, the invention is not to be restricted or limited except in accordance with the following claims and their equivalents.