Abstract:
High bandwidth, low noise switching power supply operating in quasi-resonant mode for filtering switching harmonic noise, while providing a fast control bandwidth and power at high efficiency. The power supply has an LCL tank defining a resonance period T tank , and a switching circuit regulation loop for turning on its switching circuit for on-time t ON  commencing at a time based on a state of regulation of the power supply. A switching capacitor current sensor triggers the switching circuit to turn off at the end of the resonance period T tank , whereupon the switching and output inductors enter a discharging phase for a period not related to resonance period T tank . “Quasi-resonant” operation, where the power supply is in resonant mode during on time t ON  and not in resonant mode during off time t OFF  ensures that the output inductor filters the switching harmonic noise of the switching circuit.

Description:
RELATED APPLICATIONS 
       [0001]    This application is related to U.S. application entitled “Dynamic Power Supply Employing a Linear Driver and a Switching Regulator” filed on the same date. 
       FIELD OF THE INVENTION 
       [0002]    This invention relates generally to power supplies deploying switching regulators with high bandwidth and low noise, and more particularly to switching power supplies that filter switching harmonic noise. 
       BACKGROUND ART 
       [0003]    Many electronic devices tend to require a highly responsive, low noise, high efficiency switching regulator. 
         [0004]    Switching regulators emit output current ripple at their switching frequency. That current is converted to switching voltage noise when driving a resistive load. This type of noise can be disruptive to systems that amplify low level signals, for example, where the switching noise can couple into the circuitry and become amplified along with the desired signal. Harmonics of the switching frequency, in particular can generate noise into radio bands and produce spurious transmissions or desensitize receivers. 
         [0005]    One option is to add filtering at the output of the switching regulator. Common L-C low-pass filters can be used. Unfortunately, filtering also limits the responsiveness of the switching regulator to abrupt changes in load, line, or voltage setting. Since the filters exhibit delay, it is generally not possible to include these filters inside the switching regulator&#39;s control loop for stability reasons. Thus, low noise and responsiveness are frequently tradeoffs against one another. 
         [0006]    The switching frequency of the switching regulator could be increased, allowing a higher cutoff frequency filter. However, higher switching frequencies result in higher magnetics losses and higher switching losses, decreasing power conversion efficiency. 
         [0007]    Therefore, there remains a need for a power supply that remains responsive, low noise, and maintains high efficiency. 
       SUMMARY OF THE INVENTION 
       [0008]    The objects and advantages of the invention are secured by a switching power supply that operates in a quasi-resonant mode and incorporates substantial output filtering for low switching harmonic noise, while providing a fast control bandwidth and power at high efficiency. More precisely, the low-noise, high bandwidth switching power supply is designed to operate in the quasi-resonant mode for filtering switching harmonic noise produced by a switching circuit. The switching power supply has an LCL (inductor-capacitor-inductor) tank made up of a switching inductor, a switching capacitor and an output inductor. These three elements determine a resonance period T tank  in accordance with the rules of LC-circuits. Further, the switching circuit has a control input and a switching output that is connected in series with the switching inductor. 
         [0009]    The switching power supply has a switching circuit regulation loop for turning on the switching circuit for a certain on-time t ON . The on-time t ON  starts at a time that is based on a state of regulation of the switching power supply. 
         [0010]    Once on, the switching power supply uses a switching capacitor current sensor for triggering the switching circuit to turn off at the end of the resonance period T tank . This means that the on-time t ON  is substantially equal to the resonance period T t . Specifically, the switching capacitor current sensor triggers when the switching capacitor current reaches zero after a full resonance period T tank . Then, during an off time t OFF , the switching inductor and the output inductor enter a discharging phase for a period not related to resonance period T tank . Rather, the discharging phase lasts for a period defined by off time t OFF  that ends with the re-initiation of the on-period or on time t ON . The re-initiation of the on-period is based on a state of regulation of the switching power supply as just described. Thus, a “quasi-resonant” mode of operation is enabled where the switching power supply is in resonant mode during on time t ON  and not in resonant mode during discharging phase lasting for the duration of off time t OFF . 
         [0011]    The output inductor provides filtering of the switching harmonic noise or ripple generated by the switching circuit. It should be noted that the termination of the on-off switching cycle according to the invention does not necessarily take place at either “zero current” or “zero voltage” across the switching circuit. 
         [0012]    Preferably, the switching power supply uses a pair of complementary switches in its switching circuit for switching between the on-period (for on-time t ON ) and off-period (for off-time t OFF ). In fact, this pair constitutes the “switch” of the switching circuit. 
         [0013]    The monitoring of the state of regulation of the switching power supply can be implemented in different ways. For example, the state of regulation can be determined from a certain current level of a switching inductor current. In this case, a comparator is provided for comparing the switching inductor current with a set current i SET  that corresponds to the predetermined level at which the switching circuit is to be turned on. 
         [0014]    In another embodiment, the state of regulation of the switching power supply is determined from an output voltage of the switching power supply. Here, an output voltage sensor is provided in the switching circuit regulation loop for monitoring the output voltage of the switching power supply. 
         [0015]    In some embodiments the switching power supply is further connected with a linear driver by a combining network. Such network has a node for summing an output of the linear driver with an output of the switching power supply. When the switching power supply operates in a current mode the outputs are currents. On the other hand, when switching power supply operates in a voltage mode, the outputs are voltages. 
         [0016]    The combining network has a driver feedback loop for the linear driver. In the preferred embodiment, the driver feedback loop has a capacitor coupled in series with the linear driver&#39;s output. As a result, the capacitor voltage V C  corresponds to an integral of the current components contained in the driver&#39;s output and may be used in the combining network for improving feedback control of the switching power supply. 
         [0017]    The invention further extends to a method for filtering switching harmonic noise in the switching power supply operating in quasi-resonant mode. The method calls for providing the LCL tank whose resonance period T tank  is determined by the switching inductor, switching capacitor and output inductor. The switching output of the switching circuit is connected in series with the switching inductor of the LCL tank. Furthermore, the switching circuit regulation loop is provided for turning on the switching circuit for on-time t ON , starting at the time determined by the state of regulation of the power supply, for example, when the switching inductor current reaches a certain level or when the output voltage of the switching power supply reaches a certain value. The end of the resonance period T tank  is determined from the switching capacitor current by using a switching capacitor current sensor. In fact, the end of resonance period T tank  coincides with zero switching capacitor current. 
         [0018]    The continued discharge of the output inductor during the off-time t OFF  in quasi-resonant mode results in the desired filtering of the switching harmonic noise. It should be noted that the method can be implemented in a current mode or in a voltage mode, depending on desired application of the switching power supply and type of load. 
         [0019]    Furthermore, the method can be combined with methods deploying a linear driver and combining the outputs of the linear driver and the switching power supply. 
         [0020]    Clearly, the apparatus and method of invention find many advantageous embodiments. The details of the invention, including its preferred embodiments, are presented in the below detailed description with reference to the appended drawing figures. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWING FIGURES 
         [0021]      FIG. 1  is a diagram of a switching power supply according to the invention. 
           [0022]      FIG. 2  is a detailed diagram of the switching circuit belonging to the switching power supply of  FIG. 1 . 
           [0023]      FIG. 3  are plots of relevant current and voltage waveforms in the switching power supply of  FIG. 1 , when operating in a quasi-resonant mode according to the invention. 
           [0024]      FIG. 4  is a diagram of a switching power supply according to the invention in which the state of regulation of the switching power supply is determined from an output voltage level. 
           [0025]      FIG. 5  is a diagram of another switching power supply according to the invention designed to measure switching capacitor current efficiently. 
           [0026]      FIG. 6  is a diagram of a power supply using a switching power supply according to the invention with a linear driver and a combining network for adding the outputs of the switching power supply and the linear driver. 
           [0027]      FIG. 7  illustrates a load step response of a switching power supply according to the invention. 
       
    
    
     DETAILED DESCRIPTION 
       [0028]    The figures and the following description relate to preferred embodiments of the present invention by way of illustration only. It should be noted that from the following discussion, alternative embodiments of the structures and methods disclosed herein will be readily recognized as viable alternatives that may be employed without departing from the principles of the claimed invention. 
         [0029]    Reference will now be made in detail to several embodiments of the present invention(s), examples of which are illustrated in the accompanying figures. It is noted that wherever practicable similar or like reference numbers may be used in the figures and may indicate similar or like functionality. The figures depict embodiments of the present invention for purposes of illustration only. One skilled in the art will readily recognize from the following description that alternative embodiments of the structures and methods illustrated herein may be employed without departing from the principles of the invention described herein. 
         [0030]    The present invention will be best understood by first reviewing a diagram of a switching power supply  100  as shown in  FIG. 1 . Switching power supply  100  has a switching circuit  102  connected at its first input  104  to a voltage source  106 . In the present case, voltage source  106  is a battery providing a battery voltage V BATT . Furthermore, in the specific example used herein, V BATT =3.4 V. A second input  108  of switching circuit  102  is connected to ground  110 . A control input  112  of switching circuit  102  is connected to a regulation circuit  114 . 
         [0031]    Switching circuit  102  has a switch  116 , presented schematically in this figure and shown in detail below (see  FIG. 2 ). Switch  116  switches between V BATT  and ground  110  in response to a control signal  118  applied at control input  112  of switching circuit  102  by regulation circuit  114 . Switch  116  is connected to a switching output  120  of switching circuit  102 . Switching output  120  is connected in series with a switching inductor  122  having an inductance L SW . In the present example, L SW =2.2 μH. 
         [0032]    Switching power supply  100  has an LCL (inductor-capacitor-inductor) tank  124 . LCL tank  124  is made up of switching inductor  122 , a switching capacitor  126  having a capacitance C SW  and an output inductor  128  having an inductance L OUT . In this example, C SW =4.7 nF and L OUT =4.7 μH. These three elements, namely switching inductor  122 , switching capacitor  126  and output inductor  128  determine a resonance period T tank  of LCL tank  124  in accordance with the rules of LC-circuits. 
         [0033]    In particular, the resonance of LCL tank  124  has a resonance period T tank  described by the following equation: 
         [0000]        T   tank =2π√{square root over ( L   eq   C   SW )},  (Eq. 1)
 
         [0000]    where L eg  is the parallel equivalent inductance of both inductors  122 ,  128 . The equivalent inductance L eq  is given by: 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       L 
                       eq 
                     
                     = 
                     
                       
                         
                           L 
                           SW 
                         
                          
                         
                           L 
                           OUT 
                         
                       
                       
                         
                           L 
                           SW 
                         
                         + 
                         
                           L 
                           OUT 
                         
                       
                     
                   
                   , 
                 
               
               
                 
                   ( 
                   
                     Eq 
                     . 
                     
                         
                     
                      
                     2 
                   
                   ) 
                 
               
             
           
         
       
     
         [0000]    and is equal to approximately 1.5 μH in the present example. Thus, given the exemplary inductance and capacitance values, T tank =527 ns. Switching power supply  100  has a switching circuit regulation loop  130 . Regulation loop  130  has a capacitor current sensor  132  and also includes regulation circuit  114 . Capacitor current sensor  132  is designed to monitor a switching capacitor current i CSW . 
         [0034]    Regulation loop  130  is designed to turn switching circuit  102  on for an on-time t ON  that substantially corresponds in duration to one resonance period T tank . The initiation of on-time t ON  is set based on a state of regulation of switching power supply  100 . In the preset embodiment, the state of regulation corresponds to a current level in switching inductor  122 . 
         [0035]    An inductor current sensor  134  is provided in switching circuit regulation loop  130 . Sensor  134  measures the current in switching inductor  122  to ascertain the state of regulation of switching power supply  100  when switch  116  is connected to ground  110  during an off-time t OFF . In other words, sensor  134  measures switching inductor current i LSW  (see waveform  190  in  FIG. 3 ) when switching circuit  102  is off. It should be noted that inductor current sensor  134  could be located at any suitable position in which it could reliably measure switching inductor current i LSW , e.g., it could also be located at switching output  120  or at the other side of switching inductor  122  from switching output  120 . 
         [0036]    Switching power supply  100  has an error comparator  136  in its regulation loop  130 . Comparator  136  is connected with its inverting input  138  to sensor  134 . A non-inverting input  140  of comparator  136  is connected to a source (not shown) that provides a set current i SET  to which switching inductor current i LSW  is compared to determine the state of regulation of switching power supply  100 . An output  142  of comparator  136  is connected to regulation circuit  114 , which is also a part of regulation loop  130 . Note that output  142  will turn on when switching inductor current i LSW  drops below set current i SET . 
         [0037]    In practice, current sensor  134  may convert the sensed current to a voltage, utilizing a simple current-sense amplifier well known in the art. In other words, i SET  may be expressed by a voltage representative of the set current, and comparator  136  is then a simple voltage comparator. 
         [0038]    Meanwhile, switching capacitor current sensor  132  is connected to another comparator  144  at its non-inverting input  146 . Comparator  144  is a zero-crossing comparator. Hence, its second input, namely inverting input  148 , is simply connected to ground  110 . An output  150  of comparator  144  is connected to regulation circuit  114 . Current sensor  132  and comparator  144  may operate in a fashion similar to current sensor  134  and comparator  136 , namely by comparing voltage values representative of the current to be measured. 
         [0039]    Regulation circuit  114  has AND gates  152 ,  154  as well as a D flip-flop  156 . Inputs  158 ,  160  of gates  152 ,  154  are connected to comparators  136 ,  144 , respectively. A Q output  163  of D flip-flop  156  is connected to control input  112  of switching circuit  102  for providing control signal  118  to turn switch  116  on and off. A high level at control input  112  turns switch  116  on by connecting V BATT  at first input  104  to switching output  120 . In contrast, a low level at control input  112  turns switch  116  off by connecting switching output  120  to ground  110  via second input  108 . 
         [0040]    We now refer to  FIG. 2  to review in more detail switching circuit  102  belonging to switching power supply  100  illustrated in  FIG. 1 . Switching power supply  100  uses a pair of complementary switches, namely a P-channel metal-oxide-semiconductor field-effect transistor (MOSFET)  168  and an N-channel MOSFET  170  in switching circuit  102  to serve as switch  116 . In this arrangement, during on-time t ON , or in the on-state, transistor  168  conducts from first input  104  at voltage V BATT  to switching output  120 . During on-time t ON , transistor  170  is turned off. On the other hand, in the off-state or during off-time t OFF , it is transistor  170  that conducts from second input  108  at ground (zero voltage) to switching output  120 . Transistor  168  is turned off during off-time t OFF . 
         [0041]    The on-off switching of switch  116  between transistors  168 ,  170  is controlled by control signal  118  (see  FIG. 1 ) arriving at control input  112 . A set of gates  172  and time delay elements  174  ensure a robust, non-overlapping operation of switch  116 . Thus, only one of transistors  168 ,  170  or complementary switches is on at a time while the other is off safely before turning its complement on. 
         [0042]    Returning to  FIG. 1 , we note that output of switching power supply  100  is delivered to an output  194 . Output  194  is in series with output inductor  128 . Thus, output inductor current i LOUT  corresponds to output current i OUT  of power supply  100  in the present embodiment (i LOUT =i OUT ). A useful load, not shown, can be connected to output  194  to take advantage of power supply  100 . 
         [0043]    The operation of switching power supply  100  will now be explained by making additional reference to  FIG. 3 . The latter shows plots of relevant current and voltage waveforms, when operating in a quasi-resonant mode. 
         [0044]    According to the invention, switching circuit regulation loop  130  turns on switching circuit  102  based on the state of regulation of switching power supply  100 . In the present case, this occurs when a certain current level equal to i SET , is reached in switching inductor  122  by switching inductor current i LSW . More precisely, prior to turning switching circuit  102  on, current sensor  134  is measuring switching inductor current i LSW  and comparing it with i SET  with the aid of error comparator  136 . Once switching inductor current i LSW  drops below i SET , output  142  of comparator  136  changes from low to high output level and communicates its high state to regulation circuit  114 . 
         [0045]    Regulation circuit  114  works in the following way. Input  164  of AND gate  152  is connected to the  Q  output  162  of D flip-flop  156 . Input  166  of AND gate  154  is connected to the Q output  163  of D flip-flop  156 . The Q output  163  also connects to control input  112  of switching circuit  102  and thus provides control signal  118 . 
         [0046]    When D flip-flop  156  is in the high state then Q output  163  is high. This means that control signal  118  is high, and thus control input  112  commands switch  116  to be “on”. In this state, switch  116  connects V BATT  provided by voltage source  106  to switching output  120  via P-channel MOSFET  168 , and N-channel MOSFET  170  is turned off (see  FIG. 2 ). Thus, the on-period or on-time t ON  begins. 
         [0047]    During this on-period, regulation circuit  114  passes the output of comparator  144  to AND gate  154  via input  160 . At the same time, input  166  of AND gate  154  is already high, since it is connected to the Q output  163  of D flip-flop  156 . Now, comparator  144  is configured to detect a zero crossing of switching capacitor current i CSW , which indicates the end of a resonant period T tank  and hence the end of on-time t ON . While waiting for the zero crossing, the output of comparator  136  through AND gate  152  is blocked. 
         [0048]    Thus, during the “on” period, or during on time t ON , a positive-going current zero crossing detected at comparator  144  sends a high level to input  160  of AND gate  154 . As input  166  of AND gate  154  is also high, the high level is passed through to the CLK input (indicated symbolically by “&gt;” in  FIG. 1 ) of D flip-flop  156 . Since the  Q  output  162  is connected to the D input of D flip-flop  156 , a positive-edge CLK signal inverts the state of flip-flop  156 , and thus its output  162  goes low, resulting in control signal  118  going low and turning switch  116  off. 
         [0049]    Now switching power supply  100  is in the “off” period, D flip-flop  156  is in the low state and the Q output  163  is low, and control input  112  keeps switch  116  off. In this state, switch  116  connects switching output  120  to ground via N-channel MOSFET  170 , and P-channel MOSFET transistor  168  is turned off. Thus, the off-period or off-time t OFF  begins. 
         [0050]    Since input  164  of AND gate  152  is connected to the  Q  output  162  of D flip-flop  156 , while input  166  of AND gate  154  is connected to the Q output  163 , regulation circuit  114  passes the output of comparator  136  through AND gate  152  for detecting when the level of current in switching inductor  122  drops below i SET . This occurs while the output of comparator  144  through AND gate  154  is blocked, since input  166  is low at this time and thus the output of AND gate  154  must stay low irrespective of the level of output  150  of comparator  144 . Therefore, during the “off” period, or during off time t OFF , a high level from comparator  136  indicating i LSW  has dropped below i SET  gets passed through to the S (set) input of D flip-flip  156 . This sets the state of flip-flop  156  to high, and the Q output  163  goes high, turning switch  116  on, thus initiating another on-period. 
         [0051]    A delay  165  may be included in regulation circuit  114  to ensure that a positive-going current zero crossing detected at comparator  144  at the initiation or beginning of the “on” period does not prematurely trigger the “off” state. Instead, a high level from AND gate  154  applied to CLK input before the end of the delay time introduced by delay  165  results in no action by D flip-flop  156 . As a result, the on-period is not disrupted at its onset. 
         [0052]    Switching voltage V SW  applied at control input  112  is represented by topmost waveform  180  in  FIG. 3 , where the x-axis denotes time. A dashed line  182  indicates the start or beginning of on-time t ON . A dashed vertical line  184  denotes the end of on-time t ON . 
         [0053]    In accordance with the invention, once initiated by the attainment of current level i LSW &lt;i SET  in switching inductor  122 , the length of on-time t ON  is kept substantially equal to resonance period T tank . This is done by taking advantage of switching capacitor current sensor  132  to trigger switching circuit  102  to turn off, thereby commencing an off-time t OFF  period at the end of resonance period T tank . 
         [0054]    During resonance period T tank , current sensor  132  registers switching capacitor current i CSW , which is illustrated by waveform  186  in  FIG. 3 . In the present example, current i CSW  oscillates between about 140 mA and −140 mA during resonance period T tank . 
         [0055]    In practice, current sensor  132  may convert the sensed current i CSW  in capacitor  126  to a voltage by utilizing a simple current sense amplifier well known in the art. A current sense amplifier will typically measure the voltage across a low-valued current sense resistor placed in series with capacitor  126 . Therefore, it should be understood that i CSW  can be represented by a voltage V CSW  rather than the actual switching capacitor current i CSW . In this situation, comparator  146  is a simple voltage comparator. Note that the choice of measuring currents by corresponding voltages can also be made in current sensor  134  and comparator  136 , as previously explained. 
         [0056]    Capacitor voltage V CSW  is represented by waveform  188  in  FIG. 3 . Note that in this example, voltage V CSW  moves from a level slightly above zero, namely 0.8 V, to about 5.8 V and back to about 0.8 V as current i CSW  performs one oscillation and returns to zero. The duration of this voltage oscillation corresponds to resonance period T tank . 
         [0057]    Comparator  144  keeps comparing the voltage at input  146  to ground  110  (zero voltage) that is applied at its inverting input  148 . Once switching capacitor  126  has completed one full cycle, capacitor voltage V CSW  returns to its lowest level, namely 0.8 V and switching capacitor current i CSW  crosses zero. At this time, zero-crossing comparator  144  switches its output  150  from low to high. This change in polarity is communicated to regulation circuit  114  and switch  116  is turned off as previously described. 
         [0058]    It should be noted that on-time t ON  does not end up being exactly equal to resonance period T tank , but only substantially equal to it as follows: 
         [0000]        t   ON   =T   tank   +Δτ.   (Eq. 3)
 
         [0059]    The reason for this is Δτ, which represents a delay from switching capacitor current i CSW  zero crossing to the switching of switch  116  from on to off. Delay Δτ is due to the non-instantaneous reaction of comparator  144 , flip-flop  156  and other dead time. As an example, Δτ may be about 25 ns. 
         [0060]    The duration of off-time t OFF  corresponds to the time it takes for switching inductor current i LSW  to once again drop below i SET . At that point, the state of regulation of switching power supply  100  determined from current i LSW  being less than i SET  initiates a new cycle. As the cycle repeats, control signal  118  goes high again (see V SW  represented by waveform  180 ). Off-time, t OFF , is demarcated with hatching in both  FIG. 1  and  FIG. 3  for further clarification. 
         [0061]    Switching inductor current i LSW  is shown in  FIG. 3  with waveform  190 . The level of i SET  is also indicated. Notice that i LSW  oscillates between 495 mA and 345 mA in the present example, and that it changes discontinuously at times when control signal  118  represented by waveform  180  changes from 0 V to 3.8 V and back. Any discontinuous changes in output current i OUT  or output voltage V OUT  delivered to output  194  clearly represent undesirable noise referred to a switching harmonic noise. 
         [0062]    According to the invention, the problem of switching harmonic noise at output  194  is resolved because it is output inductor  128  that is connected to output  194 . At the beginning, during and at the end of off-time t OFF , output inductor current i LOUT , indicated by waveform  192  in  FIG. 3 , continues to decline very smoothly despite discontinuous changes of voltage V SW  from 3.8 V to 0 V and switching inductor current i LSW . Since i LOUT  corresponds to output current i OUT  of supply  100  the same is true at output  194 . Thus, any load connected to output  194  experiences as smoothly changing output current i OUT  that exhibits a low ripple. 
         [0063]    It should be remarked, that during t OFF  switching power supply  100  is not operating in a resonant mode. Instead, it is running in a quasi-resonant mode. In this mode, the characteristic response of output inductor  128  filters switching harmonic noise produced by the switching regulator. 
         [0064]    Switching power supply  100  can be operated in a continuous conduction mode. In this mode, switch  116  is being constantly switched between V BATT  and ground  110  in response to control signal  118 , and the current in switching inductor  122  never idles at zero. Here, it is important to note that the length of off-time t OFF  is determined intrinsically by loop  130  to maintain the desired output voltage V OUT . In other words, the loop adjusts a duty cycle D according to: 
         [0000]    
       
         
           
             
               
                 
                   
                     D 
                     = 
                     
                       
                         
                           V 
                           OUT 
                         
                         
                           V 
                           BATT 
                         
                       
                        
                       
                         ( 
                         
                           1 
                           + 
                           
                             
                               R 
                               loss 
                             
                             
                               R 
                               load 
                             
                           
                         
                         ) 
                       
                     
                   
                   , 
                 
               
               
                 
                   ( 
                   
                     Eq 
                     . 
                     
                         
                     
                      
                     4 
                   
                   ) 
                 
               
             
           
         
       
     
         [0000]    in which R loss  stands for loss resistance and R load  stands for load resistance. 
         [0065]    Using values V OUT =2V, V BATT =3.4V, R loss =0.26Ω and R load =5.7Ω, duty cycle D is approximately 61.5%. 
         [0066]    The overall switching period T SW  of switch  116  is defined based on on-time t ON  as: 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       T 
                       SW 
                     
                     = 
                     
                       
                         t 
                         ON 
                       
                       D 
                     
                   
                   , 
                 
               
               
                 
                   ( 
                   
                     Eq 
                     . 
                     
                         
                     
                      
                     5 
                   
                   ) 
                 
               
             
           
         
       
     
         [0000]    since 
         [0000]        t   ON   ≈T   tank . 
         [0067]    From equation 1 we have calculated that T tank =527 nsec, we can calculate that T SW =857 nsec, giving a switching frequency F SW =1/T SW =1.17 MHz. Note that this switching frequency will change according to the input and output voltages V BATT  and V OUT . 
         [0068]      FIG. 4  illustrates another switching power supply  200  according to the invention. Power supply  200  is very similar to power supply  100 , but it is designed for ascertaining the state of regulation of switching power supply  200  in a different manner. 
         [0069]    Power supply  200  deploys current sensor  134  to measure switching inductor current i LSW  and supply the result to inverting input  138  of error comparator  136 , as before. However, in this embodiment non-inverting input  140  of error comparator  136  is not connected to receive i SET . Instead, it is connected to the output of a voltage error amplifier  202 . A non-inverting input  204  of amplifier  202  is connected to a certain voltage V SET , while an inverting input  206  of amplifier  202  is connected at a node  208  to an output  210  of power supply  200 . An output capacitor  212  also called a reservoir capacitor is also connected at node  208 . Typically, capacitor  212  has a comparatively large capacitance, and in the particular example described herein it is about 2.2 μF. 
         [0070]    With these connections, power supply  200  uses an expanded switching circuit regulation loop  214  that includes all the elements of prior loop  130  as well as amplifier  202 . The output of amplifier  202  corresponds to the error voltage between an output voltage V OUT  at output  210  and the selected voltage V SET . The error voltage at the output of amplifier  202  is applied to non-inverting input  140  of error comparator  136 . The latter turns on when the error voltage increases above the voltage corresponding to i LSW  as determined by sensor  134 . 
         [0071]    The regulation state of switching power supply  200  is thus determined from the level of output voltage V OUT , and amplifier  202  serves as an output voltage sensor in switching circuit regulation loop  214 . Note that since amplifier  202  is in regulation loop  214  it also strives to minimize the difference between V OUT  and V SET . 
         [0072]      FIG. 5  illustrates a switching power supply  300  similar to power supply  200 , but designed for more efficient measurement of switching capacitor current i CSW . In this embodiment, a separate current sense capacitor  302  is connected in parallel with switching capacitor  126 . Current sense capacitor  302  has a capacitance C SW-SNS  that is significantly lower than the capacitance C SW  of switching capacitor  126  of LCL tank  124 . For example, C SW-SNS  can be on the order of 0.4 nF to 40 pF, while C SW  is 4.7 nF. 
         [0073]    A current sense resistor  304  connects capacitor  302  to ground  110 . The resistance R SW-SNS  of resistor  304  is selected such that a voltage corresponding to i CSW  can be measured between capacitor  302  and resistor  304  without drawing much power. As in the previous embodiment, the voltage representing i CSW  is connected to non-inverting input  146  of zero-crossing comparator  144 . 
         [0074]    Since capacitors  126 ,  302  are connected in parallel, their capacitances add. The current passing through sense capacitor  302  is proportional to the ratio of capacitances C SW-SNS /C SW . Thus, i CSW  can be measured with sense capacitor  302  without connecting to switching capacitor  126 . 
         [0075]    Due to its small capacitance C SW-SNS , the current actually drawn by sense capacitor  302  is much lower than i CSW . Thus, the overall V·I power losses associated with measuring i CSW  are greatly reduced in switching power supply  300 . 
         [0076]    A person skilled in the art will recognize that the addition of sense capacitor  302 , despite its small capacitance C SW-SNS , will slightly modify the equations of LCL tank  124 . In particular, resonance period T tank  will be affected. This change needs to be taken into account using the above equations to ensure proper operation of switching power supply  300 . 
         [0077]      FIG. 6  is a diagram of a power supply  400  that has a switching power supply  424  in accordance with the invention and additionally deploys a linear driver  402  to handle high frequency components of the output, here represented by voltage V OUT  at an output  404 . The combination of linear drivers with switching regulators is generally desirable because with proper interconnection, a power supply using both can efficiently provide output components ranging from DC to high frequencies, e.g., up to 20 MHz. Several advantageous ways of interconnecting linear drivers with switching regulators and the corresponding combining networks are described in more detail in U.S. patent application entitled “Dynamic Power Supply Employing a Linear Driver and a Switching Regulator” and filed on the same date as the present application. 
         [0078]    Power supply  400  has a combining network  406  for adding the outputs of switching power supply  424  and linear driver  402 . Network  406  includes a summing or combining node  408  before output  404 . Node  408  is preferably a simple wired connection. Its purpose is to sum an output of linear driver  402  with an output of switching power supply  400  and deliver them to output  404 . 
         [0079]    Combining network  406  includes a driver feedback loop  410 . Loop  410  is a negative feedback loop around output  404  and between a driver output  412  of linear driver  402  and its inverting input  414 . Loop  414  includes a capacitor  416  connected in series with driver output  412 . 
         [0080]    A non-inverting input  418  of driver  402  is designed to receive a control signal  420  that output V OUT  of entire power supply  400  is trying to produce. Control signal  420  contains a number of frequency components, ranging from DC up to about 20 MHz. 
         [0081]    Combining network  406  further includes a connection between driver  402  and switching power supply  424 . Namely, non-inverting and inverting inputs  204 ,  206  of voltage error comparator  202  are connected to capacitor  416  at driver output  412  to receive capacitor voltage V C  generated across it. Specifically, non-inverting input  204  and inverting input  206  are connected across capacitor  416  such that capacitor voltage V C  is applied between inputs  204 ,  206 . 
         [0082]    Inputs  204 ,  206  correspond to an input  422  of switching power supply  424  of power supply  400 . Switching power supply  424  is indicated in a dashed line and, as remarked above, and it is frequently referred to in the art as a switching regulator. 
         [0083]    Input  422  of switching regulator  424  is thus controlled by capacitor voltage V C , which determines regulator output current i LOUT  at output of LCL tank  124 , or, equivalently, at the output of switching regulator  424 . That is because, given the interconnection between driver  402  and switching power supply  424 , error comparator  202  will strive to minimize capacitor voltage V C  by regulating output current i OUT  of switching regulator  424 . Note that the polarity of capacitor voltage V C  will determine whether regulator output current i OUT  is positive or negative. 
         [0084]    Advantageously, power supply  400  has an offset voltage source  426 , here in the form of a digital-to-analog converter (DAC), to provide a precise DC offset voltage at input  422  of switching regulator  424 . DAC  426  is connected to a voltage source (e.g. a battery), via connection  428 . 
         [0085]    Because network  406  provides for the interconnection between driver output  412  and input  422  of switching regulator  424  in the manner described above, DAC  426  can provide the DC offset in order to set the DC level of the output of driver  402 . More precisely, regulator  424  attempts, via its output, to boost the voltage at an output capacitor  430 . It does so in order to servo the average voltage output of linear driver  402  to the DC offset voltage provided by DAC  426 . 
         [0086]    During operation of power supply  400 , switching regulator  424  regulates its output to minimize the difference between the DC and low frequency components present at driver output  412  and the offset voltage from DAC  426  utilizing its regulator feedback loop established by network  406 . In other words, switching regulator  424  attempts to counteract the offset voltage and capacitor voltage V C  of capacitor  416  at its regulator output. To accomplish this, switching regulator  424  generates at its regulator output a regulator output current that forces the DC current component of capacitor  416  to be zero. The result is that switching regulator  424  contributes DC and low frequency current components to output  404 . 
         [0087]    Meanwhile, driver  402  generates the high frequency portions of control signal  420 . In the process, because its feedback loop  410  closes around output  404 , it also removes residual ripple in the output of switching regulator  424 . Since, as seen above, the ripple is smooth and does not contain any high frequency switching noise, its removal by driver  402  is very effective. For more information about the process of removal or cancellation of the ripple from switching regulator  424  by driver  402 , the reader is referred to the teachings contained in U.S. patent application entitled “Dynamic Power Supply Employing a Linear Driver and a Switching Regulator”. 
         [0088]      FIG. 7  illustrates a load step response of a switching power supply according to the invention. Notice that switching inductor current i LSW  as well as output inductor current i LOUT  track the load step within 5 μsec. This is a very short time period and permits switching power supply according to the invention to be used even in very demanding, high-speed applications. Once again, note the lack of ringing in output inductor  128 , which is a direct result of the quasi-resonant mode of operation. 
         [0089]    In view of the above teaching, a person skilled in the art will recognize that the apparatus and method of invention can be embodied in many different ways in addition to those described without departing from the spirit of the invention. Therefore, the scope of the invention should be judged in view of the appended claims and their legal equivalents.