Abstract:
A method and multiple carrier linear amplifier circuit reduces intermodulation distortion by sampling the multiple carrier linear amplifier radio frequency signal and detecting the sampled signal at frequency increments and quantizing and nulling the intermodulation distortion. The signal can have predetermined frequency increments f o  . . . f i  situated in at least one of predetermined sub-bands. The sampled radio frequency signal can be mixed with the local oscillator signal to target the centers of the multiple carriers and generate an intermediate frequency signal.

Description:
FIELD OF THE INVENTION 
   This invention relates to the field of linear amplifiers, and more particularly, this invention relates to a method and circuit for canceling intermodulation distortion in multiple carrier linear amplifiers. 
   BACKGROUND OF THE INVENTION 
   Radio frequency amplifiers use linear amplifiers that are not always “clean,” and in operation, often produce intermodulation distortion. This distortion creates interference at the operating frequencies used by radio frequency, cellular, and other similar communications circuits. As cellular and other communication systems become more important and prevalent, it is necessary to reduce the intermodulation distortion produced in amplifier systems, and especially linear power amplifiers, which play an important part in these systems. 
   Some prior art techniques suppress intermodulation distortion by generating an inverse distortion signal and applying it to the input end of the amplifier. Another prior art technique uses a negative feedback system for negatively feeding back the distortion signals with an output signal at its input end. Other prior art techniques use a feed forward system that extracts the intermodulation distortion signals, and inverts the phase of those signals to cancel the problematic signals. This type of system is widely used in base stations, satellites, and mobile communication systems. This system has high efficiency in suppressing the signals as compared to other types of linear power amplifiers, but is typically complex in structure and large. Sometimes the circuits cause increased power losses. Decreasing intermodulation distortion becomes even more important in multiple carrier linear amplifiers that are operative with multiple carriers in communications systems. Even small amounts of intermodulation distortion can create extreme performance problems. 
   SUMMARY OF THE INVENTION 
   A method of the present invention is associated with a multiple carrier linear amplifier circuit and reduces intermodulation distortion by sampling the output of the multiple carrier linear amplifier radio frequency signal and detecting the sampled signal at frequency increments and quantizing and nulling the intermodulation distortion. The circuit can generate a local oscillator signal having predetermined frequency increments F o  . . . F i  situated in at least one of predetermined sub-bands. The sampled radio frequency signal is mixed with the local oscillator signal to target the centers of the multiple carriers and generate an intermediate frequency signal. This signal is detected and digitized for quantization and nulling of the intermodulation distortion. 
   The method can also include the step of filtering the resultant intermediate frequency signal before detecting and digitizing for quantization. The circuit is stepped for the local oscillator frequency increments F o  . . . F i  and the outputs of the stepping operation are compared to identify sub-bands. The circuit identifies which frequencies are active in which sub-bands, and adjusts the local oscillator frequency based on determined active frequencies. The generated frequency increments F 0  . . . F 11  can be incremented in five MHz increments. The radio frequency signal can be generated in the radio frequency range from about 2110 to about 2170 MHz. The radio frequency signal can be divided into three sub-bands, each sub-band having up to four carriers. The intermediate frequency signal can be detected within a sample and hold circuit having a detector operative therewith. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Other objects, features and advantages of the present invention will become apparent from the detailed description of the invention which follows, when considered in light of the accompanying drawings in which: 
       FIG. 1  is a circuit of an intermodulation distortion nulling circuit that isolates amplifier output by subtracting the clean signal from the distorted output signal while using a large delay line. 
       FIG. 2  is an alternate and improved intermodulation distortion nulling circuit of the present invention where the output of the amplifier system is used to extract the required intermodulation distortion information. 
       FIG. 3  is a circuit diagram of the intermodulation distortion identification and quantization circuit shown in  FIG. 2 . 
       FIGS. 4-9  illustrate various sub-bands relative to the ACP/AACP thresholds for different carriers. 
       FIG. 10  is a high level flow chart illustrating an example of the algorithm that can be used for the intermodulation distortion nulling circuit of the present invention shown in  FIG. 2 . 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   The present invention will now be described more fully hereinafter with reference to the accompanying drawings, in which preferred embodiments of the invention are shown. This invention may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. Rather, these embodiments are provided so that this disclosure will be thorough and complete, and will fully convey the scope of the invention to those skilled in the art. Like numbers refer to like elements throughout. 
     FIG. 1  illustrates a first intermodulation distortion nulling circuit  18  as part of a multiple carrier linear amplifier (MCLA) that isolates the amplifier output intermodulation distortion and subtracts the clean input signal from the distorted output signal, thus retrieving the intermodulation distortion products. For purposes of description, the general connections among components is first described, followed by a brief working of the circuit. Further details of this type of circuit are described in U.S. patent application Ser. No. 09/564,321 filed May 3, 2000, as docket number Hoffmann 2, the disclosure which is hereby incorporated by reference in its entirety. 
   The circuit works with a carrier cancellation loop. The circuit shown in  FIG. 1  uses a large delay line, DelayS, to achieve linear operation across a wide frequency band application. This delay line significantly increases the weight and cost of a multi-channel, i.e., multiple carrier, linear amplifier. In this circuit, gain A and gain B amplifiers are balanced amplifiers. An output is sampled and the signal injected as inputs to gain A and gain B amplifiers. There are two illustrated delays in the circuit shown in  FIG. 1 , delay N and delay B. Because the amplifiers are wideband, any delay compensates for the amplifiers. If the amplifiers are identical, then the delays are identical. 
   For purposes of the description of  FIG. 1 , the interconnection among various circuit components are described, followed by their function. Gain A amplifier  20  and Gain B amplifier  21  are balanced amplifiers. Gain A amplifier  20  connects to couplers DC 1   22  and DC 2   24  and to delay B circuit  26 , which series connects to couplers DC 4   28  and DC 9   30 . A radio frequency signal  32  enters through an attenuator  34  into the coupler DC 8   36  and into coupler DC 1   22 . The delay line, DelayS  38 , is coupled from coupler DC 8   36  and connects to coupler DC 10   40 . From coupler DC 1   22 , the signal passes to the gain A amplifier  20  into coupler DC 2   24 , as noted before. Series connected from the coupler DC 4   28  is the signal combination control circuitry  39   a , including diode detector  39 , analog-to-digital converter  40 , the power null circuit  42 , and the digital-to-analog converter  44 , which then passes signals to the AttB circuit  46 , the Phase B circuit  48 , coupler DC 3   49  and Gain B amplifier circuit  21 . Coupler DC 10   40  also receives input signals via a delay line connected to coupler DC 8   36 , and coupler DC 10   40 . The signal passes from DC 10   40  as a detected signal into the distortion cancellation control circuitry  50   a  having a diode D 3   50 , analog-to-digital converter  52 , intermodulation distortion null circuit  54 , (IMD null), the digital-to-analog converter  56  and into the multiple carrier linear amplifier circuit segment shown generally by dotted line  58 . 
   The coupler DC 1   22  is connected into the delay A circuit  60  and into adjustment circuit  61 . In this embodiment, the adjustment circuit is an independent adjustment circuit  61  where phase and/or amplified signal components are independent. In other embodiments, the phase and/or amplified signal components can be adjusted together. Thus, the invention can be accomplished independent or dependent (together). The circuit  61  includes coupler DC 5   62 , Delay N circuit  64 , AttB circuit  46  and Phase delay B circuit  48 , in series. The carrier null circuit  68 , as illustrated, includes a digital-to-analog converter circuit  70  and an analog-to-digital converter circuit  72  with DAC  70  connected to AttF circuit  74  and Phase F circuit  76 , coupling to DC 5   62  and coupler DC 6   78 , which, in turn, connects to coupler DC 2   24 . Coupler DC 7   80  is ADC  72  connected and also connects to coupler DC 6   78 , diode  81 , and to 180 degree phase delay  82   a  and phase shifter N circuit  82 , and series connected linear noise amplifier  84 , and AttN circuit  86  and DC 3 . 
   It should also be understood that the entire carrier cancellation line shown at  62 ,  74 ,  76  can be eliminated. The carrier and distortion would be adjusted at the coupler  89 . It is then possible to have an adjuster as the coupler at  89 , which adjusts the phase and/or gain on both, i.e., the carrier and distortion. Thus, it is possible that the circuit could be used in other locations to detect other circuit functions, for example, to detect the carrier signal and the active sub-bands at the carrier null circuit  68 . 
     FIG. 1  shows the general block diagram of the amplifier architecture or system  18 , and includes a first amplifier path  87  and a second amplifier path  88  carrying replicas of signal components. On the first amplifier path  87 , the first amplifier  20  amplifies signal components and generates distortion components. A replica of the amplified signal components and distortion is provided to a coupling path  89 . The adjustment circuit  61  receives the distortion components from the coupling path  89  and the signal components from the second path  88  to independently adjust the phase and/or gain of at least one of the signal components and the distortion components, which adjusts the gain and/or phase relationship between the signal components and the distortion components. In this embodiment, the adjustment circuit  61  isolates the distortion components on the coupling path by combining signal components from the second path  88  and the signal components on the coupling path  89 , which are about 180 degrees out of phase and substantially equal in amplitude and thus canceled. The distortion components are amplitude and/or phase adjusted by the phase shifter  82  and the attenuator  86 . Because the signal components have been substantially removed from the coupling path, the phase and/or gain adjustments to the distortion components are made without a corresponding adjustment to the phase and/or amplitude of the signal components. 
   The adjusted distortion components are coupled onto the second path where the signal components and the adjusted distortion components are amplified by the second amplifier  21 . The amplified signal components and distortion components on the second path  88  are combined with the amplified signal components and distortion components on the first path  87  to combine constructively the signal components and destructively combine the distortion components. In the embodiment of  FIG. 1 , using the illustrated independent adjustment circuit (although the circuit does not have to be independently adjusted), when the distortion components are adjusted relative to the signal components, the phase and/or gain relationship between the signal components and the distortion components becomes independent. Thus, phase and/or gain adjustments to the distortion and signal components can be made, which improve both the constructive combination of the signal components and the destructive combination of the distortion components. 
   In operation, the amplifier system  18 , the coupler  36 , such as a 10 dB coupler (DC  8 ), receives the signal RFin and couples replicas of the signal Rfin  32  onto the first amplifier path  87  and the second amplifier path  88  after an initial amplitude adjustment of RFin by the attenuator (AttIn)  34 . The coupler provides the signal components on the first path  87  with 0 degrees phase shift and 10 dB of attenuation. The signal components are provided to the second path  88  with little attenuation and 90 degree of phase shift delay. The amplifier  20  amplifies the signal components on the first path by Gain A to produce the amplified signal components along with distortion components generated by the amplifier with 0 degrees of relative phase shift. The coupler  24 , such as a 40 dB directional coupler, couples the signal components and the distortion components onto the first path  88  and the coupling path  89 . Using a 40 dB coupler, the signal components and the distortion components are coupled onto the coupling path with 40 dB of attenuation with no phase shift. The signal components and the distortion components remaining on the first path are delayed by a phase shift of 90 degrees with little attenuation to a phase value of −90 degrees. Further details of this type of circuit operation can be found in the incorporated by reference Hoffmann 2 patent application, U.S. patent application Ser. No. 09/564,321. 
   Distortion components isolated on the coupling path  89  are provided to a 180 degree phase delay  82   a , giving the distortion components on the coupling path a phase value of 0 degrees (−180−180=−360=0 degrees). The phase shifter  82  provides a phase adjustment to the distortion components, which is not provided to the signal components which have been substantially canceled, reduced or removed from the coupling path  89 . In this embodiment, the amplifier  84 , such as a low noise amplifier, amplifies the distortion components on the coupling path  89  by 26 dB. The attenuator  86  provides an amplitude adjustment to the distortion components which is not provided to the signal components which have been removed from the coupling path  89 . As such, the distortion components are phase and/or amplitude-adjusted independent of the signal components which have been substantially canceled, reduced or removed from the coupling path prior to the distortion components being combined with signal components on the second path  88 . By independently phase and/or amplitude adjusting the distortion components on the coupling path  89 , the destructive combination of the corresponding distortion components at the output of the amplifier architecture  18  can be independently controlled and improved. 
   In this circuit, in addition to making the relative gain and/or phase adjustments between the distortion components independent of the relative phase and/or gain adjustments to the signal components, the signal components on the first path  87  become independent of the signal components on the second path  88 . 
   In other configurations where the power of the signal components is distributed among first and second amplifier paths, equal power at the inputs to first and second amplifiers on the separate paths can be achieved by sampling the output of the first amplifier, rotating the phase of the sample, and attenuatively adding the sample to the signal components on the second path to reduce the level of the signal components through what can be referred to as vector attenuation. As such, the signal components input to the second amplifier are dependent upon the output to the first amplifier. 
   The system  13  also distributes the power of the input signal components on the first and second amplifier paths  87 , 88 , thereby enabling improved power efficiency. In this circuit, however, the signal components on the first path  87  are independent from the signal components on the second path  88 , for example, by passively coupling and attenuating the signal components on the second path without vector attenuation. Because the signal components are removed from the coupling path  89 , the signal components on the second path  88  provided to the second amplifier  21  (GainB) are independent of the signal components output from the amplifier  20  (GainA) on the first path  87 , in that the amplified signal components from the first amplifier  20  will not affect the signal components on the second path  88 . Additionally, the loss of the first amplifier  20  (GainA) will not result in an undesired large increase in power level at the combined output of the system  18 . Instead, about one half of the power of the signal components would be produced. 
   The adjusted distortion components on the coupling path  89  are provided to the coupler  49 , such as a 10 dB directional coupler, which attenuates the distortion components on the coupling path  89  by about 10 dB and combines the distortion components from the coupling path with the signal components on the second path  88 . Before being provided to the coupler  49 , the signal components from the coupler are delayed by the delay  64  (DelayN) by an amount such that the distortion components on the coupling path arrive at the coupler at substantially the same time as the signal components corresponding to the distortion components. The signal components corresponding to the distortion components are the signal components which resulted in the distortion components when the signal components were amplified. The attenuator  46  adjusts the amplitude of the signal components on the second path  88 . A phase delay  48 , such as a 90 degree phase delay, delays the signal components on the second path  88  by 90 degrees to have a phase value of −90 degrees. The attenuator  46  and the phase delay  48  provide gain and phase adjustments to the signal components on the second path  88  without a corresponding change to the distortion components and thereby could be considered as part of an independent adjustment arrangement. The delay  64 , the attenuator  46  and the phase delay  48  provide constant time, amplitude and phase adjustments to enable the different paths carrying components to be combined to match up in terms of time, gain and phase for improved combining given the components used in this embodiment. 
   The signal components on the second path  88  at −90 degrees and the adjusted distortion components on the coupling path  89  at 0 degrees are provided to the coupler  49 . In this embodiment, the coupler  49  phase shifts the signal components on the second path  88  by 90 degrees to about −180 degrees and combines the signal components with the distortion components from the coupling path at about 0 degrees onto the second path. As such, the signal components with phase values at about −180 degrees and the distortion components with phase values at about 0 degrees are provided onto the second path in this embodiment. However, the 180 degree out of phase relationship and/or the amplitude difference between the signal components and the distortion components on the second path  88  can be changed due to the independent adjusting of the phase and/or amplitude of the distortion components on the coupling path  89 . 
   An attenuator  46  could adjust amplitude and the phase shifter  48  could shift the phase of the signal and distortion components. The signal and distortion components are amplified by the amplifier  21 , and the amplified signal and distortion components are combined at the coupler  30 , such as a 3 dB coupler, with the corresponding signal and distortion components on the first path  87 . The amplifier  21  amplifies the distortion components received from the second path  88  at about 0 degrees and generates distortion components at about −180 degrees from amplifying the signal components from the second path  88  which are at −180 degrees. In this circuit, the sampled distortion components from the amplifier  20  amplified by the amplifier  21  at about 0 degrees are reduced by the distortion components generated at the amplifier  21  at about −180 degrees from amplifying the signal components at −180 degrees, leaving distortion components at about zero degrees. 
   In this circuit, the signal components at the input to the amplifier  21  should have the same amplitude as the signal components at the amplifier  20  with a phase value of −180 degrees. The signal and distortion components from the coupler  28  at phase values of −90 degrees are provided to the delay  26  (DelayB) which delays the signal components and the distortion components on the first path  87  such that the corresponding portions of the signal and distortion components on the first path  87  and the signal and distortion components on the second path  88  reach the coupler  30  at substantially the same time. The amplified signal and distortion components on the first path  87  are received by the coupler  30 , which delays the signal and distortion components by 90 degrees to phase values of about −180 degrees. In producing the amplified signal components RFout, the coupler  30  constructively combines the signal components from the first and second paths  87 , 88  in phase and at about the same amplitude such that the first and second paths each provide one-half of the power to the signal components at the output of the system. Since the distortion components on the first and second paths  87 , 88  are at about 180 degrees out of phase, the distortion components on the first path destructively combine with the distortion components on the second path to reduce the distortion components at the output of the coupler  30 . 
   As noted before, the independent adjustment circuit  61  enables the relative phase and/or gain between the distortion components on the first and second paths  87 , 88  to be adjusted independent of the relative phase and/or gain adjustments between the signal components on the first and second paths. It should be understood, however, as noted before, that the adjustment circuit  61  does not have to be independent but the phase and/or amplified signal components can be adjusted together. As such, the destructive combining of the distortion components from the first and second paths at the coupler  28  can be improved by performing adjustments to the relative phase and/or gain of the distortion component on the coupling path. The power amplifier system can also provide adjustable phase and/or amplitude adjustments to the signal components which do not result in a corresponding phase and/or amplitude adjustments to the distortion components to provide adjustment of the signal components. 
   The adjustment of the relative gain and/or phase of the distortion components and/or the signal components can be performed once to align the power amplifier architecture on the production line, periodically (based on changing conditions or expiration of a time period), or dynamically (based on changing operating conditions or continuously). Because the constructive combination of the signal components can be made independent of the destructive combination of the distortion components, dynamic control to further improve the operation of the architecture can be provided in a relatively simple manner. 
   Coupler  78  can be used in conjunction with phase shifter  76  and attenuator  78  to improve cancellation of signal components. Dynamic control can also be provided by use of carrier null circuit  68  and DAC circuit  70  and ADC circuit  72 , which work in conjunction with diode detector  81  and coupler  80 . The carrier null circuit  68  acts as a power detector with the diode detector  81  to provide a power signal, indicating how well the cancellation of the signal components have been achieved. 
   Control circuitry can monitor the signal cancellation signal and provide control signals to the digital to analog (D/A) converter  70  to adjust the gain and/or phase provided by the gain  72  and phase adjusters  74  in response to the signal cancellation signal. The control circuitry provides the control signals to find the gain and/or phase adjustments, which produce a null in the cancellation signal and reflects good cancellation of the signal components on the coupling path  89 . This control can be set during initial alignment, or dynamic control provided. Dynamic control is provided because during operation any changes in the signal cancellation signal indicating a degradation in the cancellation of the signal components on the coupling path  89  can be responded to with a control signal to adjust the gain and/or phase to improve cancellation of the signal components. 
   By achieving improved cancellation of the signal components on the coupling path, the distortion components can be isolated on the coupling path, and the distortion components can be independently adjusted to improve the cancellation of the distortion components at the output of the coupler  28 . By providing for adjustment of the distortion components, control of the combination of the distortion components is possible, and dynamic control of the cancellation of the distortion components can be readily achieved, which in the presently illustrated circuit, are independently controlled. 
   A coupler  30  couples a replica of the output signal RFout onto a distortion cancellation path  90  and provides the signal to distortion cancellation control circuitry  50   a , which provides gain and/or phase adjustment control signals to gain and/or phase adjusters  82 , 86  in response to the coupled output signal. A signal on the distortion cancellation path  90  is provided to the coupler  40 , which combines the signal on the signal cancellation path with a delayed version of the signal components coupled from the coupler  36  at the input of the architecture. The signal components from the coupler  36  are delayed such that the corresponding portions of the signal components arrive at the coupler  40  at substantially the same time. The corresponding signal components should be about 180 degrees out of phase such that the signal components are reduced and the distortion components from the signal on the distortion cancellation path can be detected by detection circuitry  50 , for example including a diode detector. 
   The detection circuitry  50  provides a distortion cancellation signal indicating the level of the distortion components remaining on the output of the coupler  28 , thereby indicating the level of the cancellation of the distortion components at the coupler  28 . The distortion cancellation signal is provided to an A/D converter  52 , which digitizes the distortion cancellation signal. The digitized distortion cancellation signal is provided to control circuitry  54 . The control circuitry  54  monitors the distortion cancellation signal and provides control signals to a digital to analog (D/A) converter  56  to adjust the gain and/or phase provided by the gain and phase adjusters in response to the distortion cancellation signal. 
   The control circuitry  54  provides the control signals to find the gain and/or phase adjustments which produce a null in the distortion cancellation signal which reflects good cancellation of the distortion components at the coupler  28 . This control can be set during initial alignment, or dynamic control provided. Dynamic control can be provided because, during operation, any changes in the distortion cancellation signal indicating a degradation in the cancellation of the distortion components at the coupler can be responded to with control signals to adjust the gain and/or phase to improve cancellation of the distortion components. 
   By providing for the adjustment of the distortion components, control over the constructive combination of the signal components at the coupler  28  is possible whereby gain and/or phase adjustments are made to the signal components (alone or together with the distortion components depending on the embodiment) depending on how the constructive combination of the signal components is performed. Dynamic control of the constructive combination of the signal components can be readily achieved. In this embodiment, a signal combination signal indicative of how well the signal components are combining in the coupler  28 , for example a signal on the isolated port of the coupler, is provided to signal combination control circuitry which provides gain and/or phase adjustment control signals to gain and/or phase adjusters  46 , 48  in response to the signal combination signal. 
   The signal combination control circuitry includes the detection circuitry  39 , for example including a diode detector, which detects the signal combination signal and provides a combination signal indicating how well the signal components combined in the coupler  28 . The combination signal is provided to an A/D converter  40 , which digitizes the combination signal, and the digitized combination signal is provided to the power null, control circuitry  42 . The control circuitry  42  monitors the combination signal and provides control signals to a digital to analog (D/A) converter  44  to adjust the gain and/or phase provided by any gain and phase adjusters  46 , 48  in response to the signal combination signal. The control circuitry  50   a  provides the control signals to find the gain and/or phase adjustments which produce a null in the combination signal which reflects good constructive combination of the signal components at the coupler. This control can be set during initial alignment, or dynamic control provided. Dynamic control is provided because, during operation, any changes in the signal combination signal indicating a degradation in the combination of the signal components at the coupler can be responded to with control signals to adjust the gain and/or phase to improve constructive combination of the signal components. 
   In operation, it is evident that a signal is sampled and the main signal is cancelled because the coupler DC 5   62  samples the input and rotates it 180°. This circuit cancels from the sample and from the amplifier. What is left is the intermodulation distortion, and it is phase shifted, attenuated, and injected into the path of the other circuit signal into the gain B amplifier  21 , which amplifies the distortion. The input power to gain B amplifier  21  is equal to the input power to gain A amplifier  20 , under most conditions. Because the gain B amplifier amplifies power as gain A, distortion will also be generated and combined with the amplified distortion. 
   As a result, the signal is combined with a resultant, which will be equal in amplitude to the original distortion from the gain A amplifier, but 180° out of phase. When both distortions are added at coupler DC 4   28 , the distortions cancel each other. As will be suggested to those skilled in the art, there is a question about the distortion left at the output for coupler DC 9   30 , which samples a combined total output signal and transfers the signal to coupler DC 10   40 . At the same time, the circuit samples the input from coupler DC 8   36  via the DelayS line  38 , which is applied to the other side of coupler DC 10   40 . Both signals will be equal in amplitude and 180° out of phase and will cancel each other. 
   What will be cancelled will be the main signal, while at the inputs, there is no distortion. What will be left after cancellation is the distortion product. Whatever power is left, the circuit will detect and digitize and send into the null circuit, which will provide adjustment as a closed loop until the null is minimized. 
   In order for this illustrated circuit to work properly, a large delay line is required, as shown with the line having DelayS  38 . The entire circuit is delayed. It is not desirable to have two different phase slopes, and thus, the delay is designed into the circuit with DelayS line  38 . If there were two different phase slopes, then it would be necessary to cancel at coupler DC 3 . If there is no cancellation, then there would be a false indication of intermodulation distortion and the circuit would not be aligned properly. It is known to those skilled in the art, however, if the delay is large, it is costly to design the circuit, and the circuit will be physically large. If the amplifier in this circuit is used in a wideband configuration, such as with four different 20 MHz CDMA carriers in side-by-side relation, there will not be adequate cancellation. 
     FIG. 2  illustrates the improved circuit of the present invention using a pilotless intermodulation and quantization circuit  100  (IMD nullification circuit). In this circuit, no indication is taken from the inputs. Only the outputs are used and no delay line is necessary. The circuit of  FIG. 2  is similar in structure and function to what is shown in  FIG. 1 , with the exception of the added quantization circuit  100  of the present invention. 
     FIG. 3  is an enlarged schematic circuit diagram of the pilotless intermodulation distortion identification and quantization circuit  100  shown in  FIG. 2 . In this description, like elements from  FIGS. 1 and 2  are described with common reference numerals. 
   In this circuit, there is a simulation of the multiple carrier linear amplifier circuit  101 , where the power comes in/out, and the drive passes into the multiple carrier linear amplifier circuit. The control voltage is the same as in  FIG. 2 , and the upper coupler  102  is an equivalent for the coupler DC 9 , shown in  FIG. 2 . 
   Extending from the DC 9  coupler equivalent  102  is an isolator  104 , which could be an optional circuit component. A synthesizer circuit  106  is coupled into a heterodyned mixer circuit  108 , which is series connected to the isolator  104  and DC 9  equivalent  102 . The synthesizer circuit  106  ensures that no leakage occurs back into the output of the amplifier system. The isolator  104  allows the power to drop, and it blocks those signals that would be prone to pass back into the sample circuit corresponding to the DC 9  equivalent  102 . The mixer circuit  108  functions similar to a small receiver. 
   The synthesizer circuit  106  includes a phase lock loop circuit  110  with phase  110   a  and voltage  110   b  circuit components, and a loop filter corresponding to the operational amplifier  112  with the capacitive feedback using capacitor  114 . The phase lock loop circuit  110  connects to an oscillator circuit  116  and coupler circuit  118  with feedback from the coupler circuit  118  for closed loop operation. A digital signal processor (DSP) circuit  120  connects to phase lock loop circuit  110  and allows intermodulation distortion and adjacent channel power (IMD/ACP) optimization control. 
   This DSP circuit  120  is a generic circuit and could be a microprocessor or other control circuit, as known to those skilled in the art. The DSP circuit  120  will determine, via an algorithm of the present invention, the frequency used to tune the synthesizer circuit, which will generate the signal to be heterodyned for the output of the multiple channel linear amplifier. This signal is an intermediate frequency (IF) and passes to a low pass filter (LPF)  124 . The intermediate frequency (IF) is at a frequency used for known standards, such as the DOCOMO/UMTS frequency plan. The low pass filter  124  eliminates any harmonics and images. 
   From the low pass filter  124 , the intermediate frequency signal passes into the sample and hold circuit  130 . The signal is received within a bandpass filter (BPF) circuit  132  and will be a sharp filter, similar to a saw filter. The signal is then detected in a radio frequency detector  134 , which could be any operable type known to those skilled in the art, but in the present example, is a log detector. The signal passes into a switch  136  and capacitor  138  that together act as a sample and hold circuit. A timer signal  140  is received from the DSP circuit  120  and drives the overall circuit. The timing is selected for best performance. 
   A stream of samples is received corresponding to a DC signal representing the segments or “chunks” of bandwidth. The DC signal will pass into the analog-to-digital converter  142  and to DSP circuit  120 , where, in accordance with the algorithm of the present invention, processing occurs and decisions are based on the DC signal level. From the DSP circuit  120 , the signal passes into a digital-to-analog conversion circuit  144  corresponding to DAC  56 , and then to the multiple channel linear amplifier for phase shift and attenuation. 
   The circuit shown in  FIG. 3  is operative based on the DOCOMO/UMTS frequency plan, where the RF frequency range is 2110 to 2170 MHz. This overall band is divided/designated into three sub-bands of 20 MHz each. Each sub-band can handle up to four carriers of 5 MHz each. The total carriers per overall band are twelve. 
   In summary of the operation, the synthesizer circuit  106  generates local oscillator (LO) frequencies, which are applied to the mixer circuit. A sampled MCLA output is applied to the RF side of the mixer circuit  108 , as described before. The LO frequencies, in this example, are 2212.5 to 2267.5 MHz, 12 frequencies at 5 MHz increments, called f0, f1 . . . f11. They are mixed with the sampled MCLA RF output. The mixing targets the centers of the 12 possible carriers, at 2112.5 to 2167.5 MHz, which are also spaced at 5 MHz apart. The result is a fixed IF frequency, fLO-fRF=100 MHz. The low pass filter will eliminate the fLO+fRF products. 
   The resulting IF signal passes through the band pass filter  132 , which is centered at 100 MHz, having a passband of ˜3 MHz. The filtered RF is applied to the log detector  134  and the sample and hold (S&amp;H) circuit  136 , as described before, where it is digitized by the analog-to-digital conversion circuit (quantization)  142 . The algorithm presented in  FIG. 10  will optimize and null the intermodulation distortion, as described below. As noted before, the algorithm is described and shown relative to the W-CDMA DOCOMO/UMTS frequency plan. There are 12 possible carriers available across the band. 
   Relative to  FIG. 10 , which will be described in detail later, the basic algorithm determines active sub-bands and based on those active sub-bands, the circuit determines where the intermodulation distortion settings will be placed. Although this depends on the sub-bands, this is an implementation specific circuit. Once the intermodulation distortion settings are set, the local oscillator frequency (LO) is set such that the intermodulation distortion settings are typically next to the active sub-bands. Then the intermodulation distortion is reduced based on those measurements at that point. 
   IS-95 PCS is also 60 MHz wide, but carrier-to-carrier spacing is 1.5 MHz, which yields a total of 48 possible carriers. In reality, only 46 are available, as two carriers are not valid because of bandwidth limitations at the band edges. IS 95 also has six designated sub-bands, three 15 MHz, and three 5 MHz each. 
   For IS-136, TDMA, the frequency plan is different. Carrier to carrier spacing is only 30 kHz, but a group of up to 15 carriers is used simultaneously per sector. This pattern yields a total band of 450 kHz per sector, or roughly 0.5 MHz. Any bandpass filter in the hardware circuit will be matched per application, as well as the number of frequency steps and perhaps some logic in the algorithm. 
   In one aspect of the present invention, as a non-limiting example, fRF=2112.5 to 2167.5 MHz, 12 carriers at 5 MHz BW each, divided into three sub-bands of 20 MHz with four carriers. (Overall RF range is 2110 to 2170 MHz.) 
   fLO=2212.5 to 2267.5 MHz, 12 frequencies at 5 MHz increments, called F0, f1 . . . f11. 
   fIF=100 MHz. 
     FIGS. 4-9  are bar charts illustrating the three sub-bands of 20 MHz each, where each sub-band handles up to four carriers of 5 MHz. The adjacent channel power (ACP) and alternate adjacent channel power (AACP) graph blocks are illustrated.  FIG. 10  illustrates a flow chart for the basic algorithm used with the circuit shown in  FIGS. 2 and 3 . 
   It is shown from the flow chart that an initial sweep is made of the various frequencies. As is described above in greater detail, the synthesizer is stepped up and there are 12 different frequencies. The data coming out of the channels will have a DC voltage that has been digitized to represent the signal strength of the power coming out of the respective channel. As shown in  FIG. 4 , the first signal carrier represents an actual carrier. Frequencies 5, 6, 8 and 9 are equivalent frequencies out of the twelve frequencies at issue. This represents the intermodulation product. 
   As shown in  FIG. 5 , the two frequency blocks are side-by-side. The spaced 5 MHz spectral signals are represented by frequencies 4, 5, 8 and 9.  FIGS. 6 and 7  illustrate two carriers that are spaced 10 and 15 MHz apart respectively, but still within the single sub-band. 
     FIGS. 8 and 9  illustrate a worse case indication having four different levels with four carriers maximum per sector.  FIG. 9  illustrates the 50 channels where the Delta P (ΔP) equals 17 dB max from “pilot only” to all 50 channels. If there is a working system and all carriers are “on”, the maximum difference the system can expect from a fully loaded carrier to the pilot is about 17°. The numbers can change from system to system. It is evident from the description that the synthesizer circuit sweeps a scheme, and it is possible to download to the processor as many algorithms as desired. 
     FIG. 10  illustrates a flow chart for the algorithm that is applicable for use with the circuits shown in  FIGS. 2 and 3 . 
   As noted before, the algorithm as shown in  FIG. 10  determines active sub-bands and based on those active sub-bands determines where the intermodulation distortion setting will be placed. After this, the local oscillator is set such that the intermodulation distortion setting would be next to the active sub-bands. The intermodulation distortion is then reduced based on the measurements at that point. Thus, based on the active sub-bands, the local oscillator is set to the intermodulation distortion. Based on these active sub-bands, it is possible to know where the intermodulation distortion is located and the IMD can be detected and cancelled. 
   For example, in the flow chart, which will be described in greater detail later, at block  204 , the twelve outputs are compared and the system scans the twelve increments. Each one is looked at based on a threshold (such as if it is above a certain decibel level). If it is above that threshold, then it is an active sub-band and a determination is made as in  206   a ,  206   b ,  206   c , whether certain sub-bands are active. If not active, then the IMD settings are retained, as at block  208 . Throughout this description, F(Lo) equals the local oscillator frequency and F(x) equals the lowest active carrier frequency within a sub-band. F(X+N) equals N frequency above the lowest, while F(low) equals the inactive carrier frequency when any three out of four carriers are active. 
   For example, if FO to F3 is active, such as at decision block  206   a , then the system determines which of all four are on. If all four carriers are on, then for example, the local oscillator frequency is set to F(X+4). The different settings for different examples are shown in the figures shown in  FIGS. 4-9 , if the four carriers are on and F(X+4) is not an active sub-band, but next to it is the intermodulation distortion and that will be reduced. Thus, the sub-bands are identified at any increment, which is above a certain threshold. That is considered an active sub-band after the process is followed through as in the flow chart of  FIG. 10 . 
   Depending on those active sub-bands, the circuit determines via the algorithm how to set the local oscillator to the IMD desired. For example, if all four sub-bands are “on,” the system determines that it is at the high end of the spectrum. If the system goes higher, it is out of band. Thus, it is necessary to set the IMD to F(X−1), i.e., the next lowest increment below the band where the highest IMD will be located. If any three carriers are on, then the system moves to the inactive carrier frequency out of the group of four and to the one that is inactive where the IMD will be located. For example, F(X), F(X+1) can be the two lowest carriers, and thus, the local oscillator is set to F(X−1). The IMD will be located adjacent to it. This is shown in  FIGS. 4 ,  5 ,  6  and  7 . Thus, depending on the active sub-band, the system places the local oscillator at the spot and that is where the IMD will be at. There are, of course, different combinations as set forth in  FIGS. 4-10 . It is desirable not to go into another carrier&#39;s band. 
   The process starts and the circuit steps the LO from F 0  to F 11 , as indicated at block  200 . The detector output is recorded, corresponding to the DC signal coming out (block  202 ) and the 12 outputs are compared to identify the sub-band (block  204 ). This occurs by determining sub-band A, sub-band B or sub-band C and determining which frequencies, such as F 0  to F 3 , are active in blocks  206   a ,  206   b  and  206   c , through appropriate decision making. If none of the frequencies are active, then the intermodulation distortion settings are retained (block  208 ). This initial sweep identifies a sub-band. It is not possible to sweep the sub-band only to determine additional frequencies out-of-band, where one would expect the adjacent channel power (ACP) to be high enough and detectable. Once it is determined that a sub-band is active, decisions are made as to the carriers that are active. It is not necessary to step once again because it is loaded in memory and a threshold is set. 
   For example, if sub-band A or B is active (block  210 ), then decisions are made to check whether all four carriers are on (power equals high) (block  212 ), any three carriers are on (block  214 ), any two carriers are on and the spacing (blocks  216 ,  218  and  220 ), or whether one carrier is on (block  222 ). If yes, then the results are shown at blocks  224 ,  226 ,  228 ,  230 ,  232  and  234 . The system tests for null (block  236 ) and adjusts the respective intermodulation distortion controls, the attenuation and phase circuits (block  238 ). The settings are saved (block  240 ), and if the null is less than the threshold (block  242 ), then the stepping procedure begins once again (block  200 ). 
   If sub-band C is active (Block  244 ), corresponding to frequencies F8 to F11, then a determination is made whether all four carriers are on (block  246 ), any three carriers are on (block  248 ), any two carriers are on with different separation (blocks  250 ,  252  or  254 ), or only one carrier is on (block  256 ). If yes, then the local oscillator frequency is adjusted as indicated at blocks  258 ,  260 ,  262 ,  264 ,  266  or  268  respectively. Then the test is made for null at block  236 . 
   It is evident from this flowchart that the digital signal processor circuit will set the control voltage and start searching. It will adjust the attenuation and phase shift until minimization occurs and there is a null. In this description, F(x) and F(x+1) corresponds to adjacent carriers while F(x+3) corresponds to 15 MHz separation. 
   Many modifications and other embodiments of the invention will come to the mind of one skilled in the art having the benefit of the teachings presented in the foregoing descriptions and the associated drawings. Therefore, it is to be understood that the invention is not to be limited to the specific embodiments disclosed, and that the modifications and embodiments are intended to be included within the scope of the dependent claims.