Abstract:
The present invention includes a balanced compact antenna, conforming to the envelope restrictions appropriate to a PC-card form factor, with maximum radiation intensity along a long axis of the card. The inventive antenna configuration employs an inductive shorting bar to match an “M”-shaped bent dipole antenna to a differential feed. The combination of horizontal cross-members and large vertical downward legs ensures radiation predominantly in a broadside direction while keeping the dimensions of the antenna sufficiently compact to fit within the PC-card envelope.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
   The present application claims the benefit of and priority to U.S. Provisional Patent Application Ser. No. 60/630,509, filed on Nov. 22, 2004, the entire disclosure of which is incorporated herein by reference. 

   FIELD OF THE INVENTION 
   The present invention is related to communications using radio frequency signals, and more particularly to an improved compact antenna having a forward-directed radiation pattern. 
   BACKGROUND 
   Radio Frequency Identification (RFID) technologies are widely used for automatic identification. A basic RFID system includes an RFID tag or transponder carrying identification data and an RFID interrogator or reader that reads and/or writes the identification data. An RFID tag typically includes a microchip for data storage and processing, and a coupling element, such as an antenna, for communication. An RFID reader operates by writing data into the tags or interrogating tags for their data through a radio-frequency (RF) interface. During interrogation, the reader forms and transmits RF waves, which are used by tags to generate response data according to information stored therein. The reader also detects reflected or backscattered signals from the tags at the same frequency, or, in the case of a chirped interrogation waveform, at a slightly different frequency. 
   RF readers can operate at a number of different frequency bands or ranges. Common low frequency ranges include 125–134 KHz and 13.56 MHz, and common high frequency or ultra-high frequency (UHF) ranges include 860–960 MHz, and 2.4–2.5 GHz. RFID systems operating at the low-frequency ranges are widely used and are inexpensive, but have the fundamental disadvantage that coupling between the reader antenna and the tag antenna is almost entirely inductive. As a consequence, the power that can be coupled to the tag falls rapidly when the distance between the reader and the tag is greater than roughly the antenna size. Since the reader antenna size is typically limited to around 1 meter, an interrogation range characterized by a maximum operable reader-tag separation in low-frequency systems is similarly limited to less than about 1 meter, with typical interrogation range for high data rate applications being even shorter (e.g., a few tens of cm). This interrogation range, although limited, still allows many useful applications, but when longer interrogation range is required, it is appropriate to consider UHF (i.e., 900 MHz or higher) systems, which allows much longer interrogation ranges, such as from about 3 to 8 meters, to be achieved. 
   Conventional RFID readers operating at the UHF frequency band around 900 MHz have been large, separately packaged devices attached to removable external antennas or integrated with an antenna. Examples of these readers include the ALR9780 and ALR 9040 readers from Alien Technology, the AR400 and SR400 devices from Matrics/Symbol, and the ITRF and IF5 readers from Intermec Inc. Relatively large handheld readers with integral antennas have also been reported, such as the IP3 and Sabre 1555 devices from Intermec Inc. 
   RFID readers have not been made in a PC Card format so that it can be integrated in handheld, portable or laptop computers to read from and write to RFID tags. It is apparent that incorporation of a RFID reader into a PCMCIA-compatible (“PC-card”) form factor will provide numerous practical advantages, since a user may then employ the PC-card-reader in any PC-card-compatible device, such as a laptop computer, or personal digital assistant (PDA), with only the addition of appropriate software. In this fashion virtually any portable computing device can be RFID-enabled. The flexibility of an RFID reader on a PC Card also allows easy integration of an intelligent long-range (ILR) system into enterprise systems and permits combination with other technologies such as bar code and wireless local area networks (LAN). The making of a PC Card RFID reader, however, presents many challenges, one of them is associated with the design of a suitable antenna. 
   SUMMARY OF THE INVENTION 
   The present invention includes a balanced compact antenna, which can be made to conform to envelope restrictions of a PC-card form factor, with maximum radiation intensity along a central axis of the antenna. The inventive antenna configuration employs an inductive shorting bar to match an “M”-shaped dipole antenna to a differential feed. The combination of horizontal cross-members and large vertical downward legs ensures radiation predominantly in directions along the central axis of the antenna, while keeping the dimensions of the antenna sufficiently compact to fit within a PC-card envelope. The antenna can be built on a substrate and comprises a pair of conductor lines formed on the substrate and an inductive shunt connected between the pair of conductor lines. The pair of conductor lines have a pair of feed portions extending from a pair input terminals, respectively, toward left and right edges of the substrate, a pair of riser portions extending a distance L from respective ends of the pair of feed portions toward a top edge of the substrate, a pair of radiating cross-members extending a distance L 1  from respective ends of the pair of riser portions toward left and right edges of the substrate, and a pair of downward leg portions extending a distance L 2  from respective ends of the pair of radiating cross members toward a lower edge of the substrate. The inductive shunt is parallel with the feed portions and extends between the pair of riser portions. In one embodiment of the present invention, the pair of conductor lines and the inductive shunt are arranged on the substrate such that the pair of conductor lines are positioned as mirror images of each other with respect to the central axis of the antenna, that the two input terminals are separated by a distance g, that the riser portions are each separated from the central axis by a distance H, that the inductive shunt is separated from the feed portions by a distance d, and that H+L+L 1 +L 2 −d−w≈λ/4, where w is an approximate linewidth of the riser portions and λ is the wavelength corresponding to a center frequency of a frequency band in which antenna  100  is designed to operate. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a schematic depiction of the envelop restrictions of a PC card form factor. 
       FIG. 2  is a diagram of a prior art meandered printed wire 2.4 GHz antenna. 
       FIG. 3  is a diagram of a prior art inverted-F antenna for 2.4 GHz PC-card compatible transmission with broadside radiation. 
       FIG. 4  is a diagram of a prior art variant of the inverted-F antenna with reduced lateral extent 
       FIG. 5  is a diagram of another prior art variant of the inverted F with lumped capacitor termination to reduce wire length. 
       FIG. 6  is a diagram of a prior art multiple-folded inverted-F antenna. 
       FIG. 7  is a diagram of a prior art balanced inverted-F antenna. 
       FIG. 8  is a diagram of a prior art variant of the balanced inverted-F antenna using lumped inductors. 
       FIG. 9  is a diagram of a bent differential inverted-F antenna. 
       FIG. 10  is a top view of a compact antenna with directed radiation pattern according to one embodiment of the present invention 
       FIG. 11  is a diagram illustrating geometrical definitions of an exemplary antenna used simulations. 
       FIG. 12  is a smith chart of simulated impedance response of the exemplary antenna. 
       FIG. 13  is a chart of simulated input return loss vs. frequency for the exemplary antenna 
       FIG. 14  is a chart of simulated input return loss of antennas having various values for the shunt inductor spacing d. 
       FIG. 15  includes Smith charts of simulated impedance responses for various values of shunt inductor spacing d. 
       FIG. 16  includes Smith charts of simulated impedance responses for various values of width parameters H and L 1 . 
       FIG. 17  is a diagram illustrating simulated direction and magnitude of surface currents in the exemplary antenna. 
       FIG. 18  is a diagram showing simulated distribution of the magnitude surface current in the exemplary antenna. 
       FIG. 19  is a diagram illustrating simulated radiation pattern on an x-y plane. 
       FIG. 20  is a diagram illustrating simulated radiation pattern on a y-z plane. 
       FIG. 21  is a diagram illustrating an exemplary circuit board layout of the exemplary antenna including discrete matching elements realized as surface-mount components. 
       FIG. 22  is a circuit schematic of the discrete matching elements. 
       FIGS. 23A–23C  are diagrams illustrating a planar Marchand balun that can be used to convert between a single-ended radio signal input and the balanced inputs to the antenna. 
       FIGS. 24A–24D  are top and cross sectional views of a printed circuit board accommodating a reader and the exemplary antenna. 
       FIG. 25  includes Smith charts of simulated impedance response of the exemplary antenna with and without the inductive matching stub  102 . 
       FIG. 26  includes Smith charts of measured impedance response of the exemplary antenna with and without the inductive matching stub  102 . 
       FIG. 27  includes Smith charts of measured impedance response of the exemplary antenna with and without the discrete matching elements. 
       FIG. 28  is a diagram illustrating configuration of a radome for enclosing the exemplary antenna. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1  illustrates a PC card RFID reader  10  inserted in a PC card slot of a computer system  12 , which can be a portable computer system such as a laptop computer, a PDA device, or the like. As shown in  FIG. 1 , reader  10  has an inside portion  14  that is enclosed by the computer system  12  and an outside portion  16  that is protruding from the computer system  12 . Reader  10  could employ removable antennas connected by, for example, a coaxial cable, or one or more integral antennas attached directly onto the protruded portion  16  of reader  10 . The latter arrangement provides significant advantages in size, convenience, and portability for the end user of the reader, but creates significant challenges for antenna design. For maximum convenience and simplicity of use, the integral antenna should be no larger than a width w of the card slot, and at most only slightly thicker than a thickness d of the card. From the point of view of manufacturing cost, an antenna that can be printed on conventional circuit board material, possibly even the same circuit board used for building the reader circuitry, should be greatly preferred over an antenna requiring any out-of-plane assembly. In many cases the integral antenna will be mechanically vulnerable during use, and therefore should not protrude excessively beyond the protruded portion  16  of the card. 
   In many handheld or portable applications of reader  10 , the near-field environment of the antenna is not well controlled. Thus, it is also very desirable that the antenna impedance be relatively insensitive to nearby metal or dielectric obstacles, so that good matching and power transfer to and from the reader will be maintained in the presence of people and common metallic objects. Finally, it is very desirable that the integral antenna should direct the majority of its radiation in a ‘forward’ direction pointing away from the computer system  12  (i.e., along the y axis in  FIG. 1 ), so that a user may rely upon the orientation of the computer system  110  as a somewhat-reliable indicator of the location of the responding RFID tags, at least in short read ranges. In real indoor environments, due to reflection and diffraction from numerous complex obstacles generally present, precise localization by pointing the computer system at the tag cannot realistically be attained when the distance between the reader and the tag is larger than a meter or two. 
   In summary, an integral compact antenna for reader  10  preferably meets the following design goals:
         The antenna should have insignificant geometric height, and preferably be printed on the same board material upon which the reader is built.   The antenna should mainly radiate in the forward direction when the reader is inserted in a computer system.   The useable frequency range should cover a frequency band for unlicensed operation in the United States under FCC regulations, such as 902 MHz–928 MHz (λ˜328 mm). Slightly different frequency bands may be needed for operation in other regulatory jurisdictions.   The antenna should attach to a PCMCIA card housing, and more preferably in the protruded portion of the card when the card is inserted in a computer system. As a non-limiting example, the protruded portion may have the following dimensions: L x =49 mm(0.15*λ), L y =36 mm(0.11*λ), L z =5 mm, as shown in  FIG. 1 .       

   Conventional antennas do not satisfy the above conditions. Among them, microstrip or ‘patch’ antennas are well-known, low-cost, versatile antennas. However, the main direction of radiation for a patch antenna is perpendicular to the plane of the patch. Patch antennas are also generally close to half of a wavelength in length in order to provide a near-resonant real load. At radio frequency, this length would significantly exceed that achievable using conventional printed-circuit board materials and configurations. A patch antenna is thus unsuitable for a PC card reader. 
   A meandered 2.4 GHz antenna disclosed by Lin, et al. and shown in  FIG. 2  may be configured to fit within an appropriate size envelope for a PC card reader. This antenna can also be scaled for acceptable impedance matching at 900 MHz, but the effects of currents flowing parallel to the x axis in successive horizontal arms of the meander line nearly cancel in far field. So, this antenna radiates ineffectively in the y-direction, and is thus unsuitable for a PC card reader. 
   In order to obtain significant radiation in the y-direction, one can start with a quarter-wave ‘monopole’ antenna over a ground plane, and then bend the main part of the dipole so that it is directed over the ground plane. A shunt inductor connected near the antenna feed can be used to compensate for the capacitive loading from the proximity of the ground plane. A well-known configuration of this type is the ‘inverted-F’ antenna described by, for example, Soras, Karaboikis, Tsachtsiris and Makios, which has a 2.4 GHz PC-card-compatible configuration, shown schematically in  FIG. 3 . Unfortunately, this antenna, if scaled for 900 MHz operations, would be too large for the PC-card form factor. Furthermore, this antenna has an ‘unbalanced’ design, in which a single-ended current flows in a radiating wire  300  connected to a feed line with reference to a large ground plane. As such, the antenna impedance is sensitive to the size and shape of the ground plane and therefore the configuration of the radio circuitry in the card and the card-mounting environment, as well as nearby dielectric or metallic objects. 
   Certain variations of the inverted-F antenna have been examined using simulation in an attempt to arrive at a 900-MHz version that could be contained within the required physical envelope. For example,  FIG. 4  shows a possible variant in which the radiating wire  300  is bent a second time to contain it within the allowed lateral spacing. However, because of the significant uncompensated vertical current from the final leg of the wire, this antenna directs most of its power towards the lower right, as shown in the figure. 
   As shown in  FIG. 5 , the length of the radiating wire  300  can also be reduced by placing a lumped inductor (not shown) at a feed point, or a lumped capacitor plate  500  at an end  301 . In this fashion the width of the antenna can be reduced to about 41 mm (λ/8). However, at this size the antenna is only 8 mm narrower than the PC-card-constrained ground plane, and thus the image of the antenna in the ground plane is about the same size as the antenna; therefore there is little radiation in the desired y-direction (that is, the ground plane image is too small). Attempts to reduce the width of the antenna to less than an eighth of a wavelength result in significant reduction in the radiation resistance of the antenna, making it very difficult to achieve a good electrical match. 
   In order to reduce the lateral extent of the inverted-F antenna at 900 MHz, additional bends may be added. For example, Kadambi, Yarasi, Sullivan and Hebron have disclosed a multiple-bend inverted-F having successive legs  601 , as shown in  FIG. 6 . However, like the antenna of Lin et al. in  FIG. 2 , the currents flowing in the successive legs  601  nearly cancel each other in the far field. 
   Furthermore, none of these inverted-F variants address the problem of unbalanced operation and consequent sensitivity of the match to ambient objects. Compact balanced implementations are even more challenging than unbalanced antennas. As balanced arms are required, more space is used. An example of balanced implementation of an inverted-F configuration has been provided by Schulteis, Waldschmidt, Sorgel and Wiesbeck and depicted in  FIG. 7 , which is much larger than allowed for PC card application at about 900 MHz. According to the authors, the antenna size can be reduced by up to 20% using lumped tuning elements. But that is still not sufficient to fit within a PC-card envelope. 
   Variations of the balanced inverted-F antenna have also been examined by simulation. In one variation, the electric antenna length is increased, a lumped inductor is added to the center of wire S 2  in  FIG. 7 , the center of wire S 1  is fed with a discrete port, and the antenna is further bent at the top, resulting in a configuration shown in  FIG. 8 . The final bends at the top is to achieve a further size reduction in order to fit the whole antenna within the desired size envelope, but they also cause horizontal currents (parallel to the x axis) to flow, partially canceling the effects of currents on the lower branch of the antenna and reducing desired radiation in the y-direction. 
   Another variation of the balanced inverted-F design, in which the antenna bend is placed past the location at which the inductive shunt is tapped, has been described in documents by Integration Associates, Inc., and is shown in  FIG. 9 . This configuration has better forward radiation properties since there is no cancellation of horizontal currents. Unfortunately, since the straight portion of the feed line extends past the tap, the width of the antenna becomes excessively large for the PC-card form factor. 
   Therefore, none of the prior art clearly discloses an antenna that can meet the demanding requirements set forth above for a compact, integral antenna attached to an RFID reader compatible with a PC-card form factor. 
     FIG. 10  illustrates a compact forward-directed antenna  100  according to one embodiment of the present invention. As shown in  FIG. 10 , antenna  100  comprises a pair of conductor lines  100   a  and  100   b . Conductor line  100   a  has a feed portion  101  extending from an input terminal A along a first direction, a riser portion  103  extending a length L from an end A 1  along a second direction to an end A 3 , a radiating cross-member  104  extending a length L 1  along a third direction, and a downward leg portion  105  extending a length L 2  from an end A 4  along a fourth direction to an end A 5 . Likewise, conductor line  100   b  has a feed portion  101  extending from an input terminal B along a fifth direction, a riser portion  103  extending a length L from an end B 1  along a sixth direction to an end B 3 , a radiating cross-member  104  extending a length L 1  along a seventh direction, and a downward leg portion  105  extending a length L 2  from an end B 4  along an eighth direction to an end B 5 . 
   The second direction is substantially perpendicular to the first direction, the third direction is substantially parallel to the first direction, and the fourth direction is substantially opposite to the second direction. Likewise, the sixth direction is substantially perpendicular to the fifth direction, the seventh direction is substantially parallel to the fifth direction, and the eighth direction is substantially opposite to the sixth direction. Also, the fifth direction is substantially opposite to the first direction and the seventh direction is substantially opposite to the third direction. In one embodiment of the present invention, as shown in  FIG. 10 , the first and third directions are along the x-direction, the second and sixth directions are along the y-direction, the fifth and seventh directions are along the negative x-direction, and the fourth and eighth directions are along the negative y-direction. 
   Antenna  100  further includes a third conductor line  102  extending a length H from a center C of conductor line  102  toward inner edges of riser portions  103  of conductor lines  100   a  and  100   b , and connecting with riser portion  103  of conductor lines  100   a  at point A 2  and with riser portion  103  of conductor line  100   b  at point B 2 . In one embodiment of the present invention, the pair of conductor lines  100   a  and  100   b  and the third conductor line  102  are arranged in a plane (e.g., the x-y plane) such that the pair of conductor lines  100   a  and  100   b  are positioned as mirror images of each other with respect to a center line (CL) axis parallel to the y-direction, that terminals A and B are separated by a distance g, that the riser portions  103  of conductor lines  100   a  and  100   b  are each parallel to and separated from the CL axis by a distance H, that the third conductor line  102  is substantially parallel to feed portions  101  and distanced from the feed portions  101  by a distance d, and that
 
 l=H+L+L 1+ L 2 −d−w≈λ/ 4  (1)
 
where w is an approximate linewidth of the riser portion  103  of conductor lines  100   a  and  100   b , λ is the wavelength corresponding to a center frequency, such as 915 MHz, of a frequency band in which antenna  100  is designed to operate, and l is a resonant length measured from the center C of conductor line  102  to either end A 5  of conductor line  100   a  or end B 5  of conductor line  100   b  along a center line (shown as dashed lines in  FIG. 10 ) in either conductor line  100   a , or conductor line  100   b , respectively.
 
   Still referring to  FIG. 10 , in one embodiment of the present invention, conductor lines,  100   a ,  100   b , and  102  are etched metal traces printed on a first side of a substrate  120 , such as an FR4 fiberglass composite substrate. A continuous metal ground plane  106  is formed on a second side opposite to the first side of substrate  120  to cover a portion of substrate  120  from the second side. An upper edge of the ground plane  106  is separated from the feed portions  101  by a distance s. Two parallel printed traces  108 , which form a pair of coplanar transmission lines separated by gap g, may be provided to on the first side of substrate  120  connect the antenna to a radio front end that employs differential input/output connections. For a radio that generates a single-ended voltage signal referenced to the ground plane  106 , conventional means can be employed to convert between the single-ended voltage output (not shown) from the radio and the balanced inputs to terminals A and B of antenna  100 . 
   Another continuous metal ground plane  106  may also be formed on the first side of the substrate  120  to cover the same portion of the substrate  120  from the first side. Conventional means of isolation can be used to isolate the metal ground plane  106  on the first side from the co-planer transmission lines  108  or the single ended-voltage output. 
   Conductor line  102  acts as a shunt inductor to a virtual ground potential present along the CL axis. The shunt inductor separates each of conductor lines  100   a  and  100   b  into two parts, a first part running from terminal A to point A 2  in riser  103  of conductor line  100   a  and from terminal B to point B 2  in riser  103  of conductor line  100   b , and a second part running from point A 2  to end A 5  in conductor line  100   a  and from point B 2  to end B 5  in conductor line  100   b . The shunt inductance associated with the shunt inductor  102  resonates with the impedance of the second parts of conductor lines  100   a  and  100   b , which impedance is capacitive because the second part of conductor line  100   a  or  100   b  has a length shorter than λ/4 according to Equation (1). Therefore a large amount of current should flow in the inductive shunt, that is, conductor line  102 . Since most of the current in antenna  100  flows through the inductive shunt  102 , the resonant length is approximately measured from the center of the shunt  102  rather than the center of the feed  101 . Thus, the resonant length l equals approximately to H+L+L 1 +L 2 −d−w, which is set to be about a quarter of the wavelength corresponding to the center frequency, as expressed in Equation (1). 
   The above features of antenna  100  ensure that maximum current density occurs near a midpoint in conductor line  102  and is oriented along the x-axis in order to radiate in the y-z plane that is perpendicular to the x-axis. The horizontal radiating cross-members  104  of conductor lines  100   a  and  100   b  also provide currents along the x-axis with resulting radiation maximizing in directions perpendicular to the x-axis. The currents in the radiating cross-members  104  of conductor lines  100   a  and  100   b  are approximately in-phase with that in the inductive shunt  102  and thus adds instead of cancels the current in conductor line  102 . The downward leg portions  105  of conductor lines  100   a  and  100   b  provide currents that approximately cancel the effects of currents flowing in the riser portions  103  of conductor lines  100   a  and  100   b , respectively. Thus, undesired radiation in directions perpendicular to the y direction is minimized. 
   In one embodiment of the present invention, ends A 5  and B 5  at which the downward legs  105  terminate are arranged to be close to the ground plane  106 , as shown in  FIG. 10 . This arrangement allows for a convenient addition of tip-loading overlap capacitance by slightly extending conductor lines  100   a  and  100   b  over the ground plane  106  so that conductor lines  100   a  and  100   b  each slightly overlaps with the ground plane  106 . Addition of lumped-element capacitive loadings at terminals A 5  and B 5  can also be made using surface-mount capacitors and via holes to the ground plane. A varactor diode in addition to or in place of the lumped-element capacitors or overlap capacitance may also be added to allow tuning of the antenna for real-time optimization of performance. 
   Simulations are performed to examine the performance of antenna  100  using geometries shown in  FIG. 11  and Table 1.  FIG. 12  illustrates simulated impedance response of antenna  100  drawn in a Smith chart, and  FIG. 13  illustrates simulated return loss vs. frequency for antenna  100 . Generally speaking, the input impedance Zin of antenna  100  can be expressed as Zin=Re(Zin)+j*Im(Zin), where Re(Zin) is the real impedance and Im(Zin) is the imaginary part of the impedance. Resonance occurs when, Im(Zin) is zero or near zero. As shown in  FIG. 12 , according to the simulations, antenna  100  exhibits an impedance of (37+j21) Ohm at 963 MHz frequency, an impedance of (49+j1.5) Ohm at 957 MHz frequency, and an impedance of (66−j22) Ohm at 951 MHz frequency. As shown in  FIG. 13 , the simulated return loss for antenna  100  has a dip indicating a resonant frequency at about 957 MHz. 
   
     
       
             
             
             
           
         
             
               TABLE 1 
             
             
                 
             
             
               Parameter 
               Value 
               Units 
             
             
                 
             
           
           
             
               Substrate 
               FR4 
                 
             
             
               Substrate thickness 
               710 
               μm 
             
             
               L 
               31 
               mm 
             
             
               H 
               11 
               mm 
             
             
               L1 
               9 
               mm 
             
             
               L2 
               34 
               mm 
             
             
               g 
               2 
               mm 
             
             
               d 
               2 
               mm 
             
             
               s 
               3 
               mm 
             
             
               w 
               1.3 
               mm 
             
             
                 
             
           
        
       
     
   
   For simplicity, a plastic radome, which is used to enclose the circuit board supporting the antenna, as discussed in more detail below, was omitted during simulation. The inclusion of the radome would shift the resonant frequency toward the center of the ISM band, i.e., the nominal 915 MHz. The depth of the resonance dip is associated with the real impedance of antenna  100  at resonance and is about 35 dB. A 10 dB impedance bandwidth of antenna  100  is about 15.62 MHz. 
   The depth and location of the dip can be adjusted by adjusting the geometry of antenna  100 . Simulations show that the gap d between the tuning stub  102  and the feed  101  influences the resonant frequency and the return loss at resonance.  FIG. 14  illustrates results of simulations done to show the effect of the gap d between the tuning stub  102  and the feed  101  on the resonant frequency and the return loss at resonance.  FIG. 15  shows the corresponding Smith charts of the simulated impedance response of antenna  100  when d is varied. 
     FIG. 16  shows Smith charts of the simulated impedance response of antenna  100  for different sets of H and L 1  values. Adjustments in the width parameters H and L 1  influence the radiation resistance of the antenna. These two parameters may be adjusted while maintaining the sum roughly constant to adjust the real impedance at resonance. As shown in  FIG. 16 , the resonant frequency increases when H increases and L 1  decreases. 
   Adjusting the length of the downward legs L 2  mainly affects the resonant frequency without changing the radiation resistance much; thus L 2  may be used to adjust the center frequency after the other parameters have been adapted for the desired bandwidth and return loss. 
     FIG. 17  illustrates simulated current distribution in antenna  100  where the current is shown as arrows whose directions indicate the directions of the current flow in various parts of antenna  100  and whose sizes are roughly proportional to the magnitude of the current.  FIG. 18  is a contour chart of the magnitude of the current density in antenna  100 . In the example shown in  FIGS. 15–18 , the maximum current density is about 54.3 A/m of linewidth. It is apparent that the current density is maximized in the inductive shunt or stub  102 , the vertical risers  103 , and the cross members  104 . The current density in the feed lines  101  is relatively low. The simulation agrees with the theory that the inductive stub  102  and the second parts of conductor lines  100   a  and  100   b  above the inductive stub  102  form a resonant circuit with a reasonably high quality factor Q, so that large reactive currents flow. The feed lines  101  see nearly real impedance and supply smaller real current, which is oppositely directed relative to the current in the stub  102 . The current density is high on the horizontal cross members  104 , promoting radiation along the y-axis. The current density on the left and right vertical risers  103  is oppositely directed and cancels in the far field, contributing to minimal radiation in directions perpendicular to the y-axis. 
     FIGS. 19 and 20  illustrate simulated radiation pattern of antenna  100 . As shown in the figures, the radiation from antenna  100  is omni-directional in the y-z plane and the radiation towards the front (+y) and back (−y) of antenna  100  is about equal according to the simulations. In practice the backward radiation in the −y direction would be of reduced significance both because of the presence of the device into which the reader card containing the antenna is inserted and the likely presence of a user of the reader card. In certain applications the backward radiation could represent a disadvantage, as at high output powers, there may be some concern for the safety of the user. Further work is required to establish whether the backward radiation from antenna  100  represents a problem. 
   Simulations are also performed to investigate the effect of changes in the linewidth w of conductor lines  100   a ,  100   b , and  102 . According to the simulations, changes in the linewidth w only weakly affect the behavior of the antenna; for example, a 30% change in linewidth induces roughly a 20% change in the impedance of the antenna at resonance. The risers  103  may be tilted as much as 10 degrees from the vertical towards the CL axis of the antenna with little effect on the impedance or gain of the antenna. 
   EXAMPLE 1 
     FIG. 21  illustrates a circuit board layout of antenna  100  according to one embodiment of the present invention. Antenna  100  in  FIG. 11  is constructed on a printed circuit board  120  using parameters given in Table 1. These parameters are chosen to provide good matching and radiation in the US industrial, scientific, and medical (ISM) band having a frequency range from 902 MHz to 928 MHz.  FIG. 21  also shows an input line  112  for receiving a single-ended radio signal and a conventional wire-wound balun  113  employing a ferrite core and bifilar winding, which is employed to provide a transition between the single-ended input line  112  and antenna  100 . Discrete matching components including a capacitor  114  inserted in input line  112 , an inductor  115  coupled between input line  112  and ground plane  106 , and a capacitor  116  coupled between terminals A and B of conductor lines  100   a  and  100   b  of antenna  100  are also provided to compensate for effects caused by imperfection of balun  113 . 
   Capacitor  114 , inductor  115 , and capacitor  116  are also employed to compensate for small changes in frequency that may result when a plastic radome is incorporated to protect antenna  110 , as discussed in more detail below. A schematic diagram of the matching elements is shown in  FIG. 22 . To provide some external tuning capability, a varactor diode can be used in place of or in series with capacitor  114 , or coupled between line  112  and ground plane  106  as appropriate to provide a shunt capacitance. 
   Referring back to  FIG. 21 , since length L 2  of the downward legs  105  primarily affects the resonant frequency with little change in the impedance of the antenna, antenna  110  may further include solder pads  117  placed in rows extending from terminals A 5  and B 5  in the negatively direction. Solder pads  117  are provided to allow for convenient increases in length L 2  by wire bonding or soldering and thus provide a second method of easily adjusting the resonant frequency to compensate for small changes resulting from the radome, manufacturing tolerances, effects of the remainder of the board and the portable device into which the board is installed, and other minor influences. 
   Instead of the wire-wound balun  113 , a planar Marchand balun can be used to transition between a single-ended signal input I to the balanced inputs A and B of antenna  100 . As shown in  FIG. 23A , the planar Marchand balun  130  comprises a first pair of transmission lines  132  and  134  and a second pair of transmission lines  136  and  138 , each transmission line being approximately a quarter wavelength in length. Transmission line  132  is connected between input B of antenna  100  and ground, transmission line  134  is connected between input A of antenna  100  and ground, and transmission lines  136  and  138  are serially connected with each other between the single-ended input I and a floating terminal F. Transmission line  136  is connected to the single-ended input I through a plurality of discrete matching components such as those shown in  FIG. 22 . Furthermore, transmission lines  136  and  138  are arranged to be close to transmission lines  132  and  134  to allow coupling of the signal fed to transmission lines  136  and  138  into transmission lines  132  and  134 . See also R. Schwindt, C. Nguyen, “Computer-Aided Analysis and Design of a Planar Multilayer Marchand Balun”, IEEE Trans. on Microwave Theory and Techniques, July 1994, vol. 42, issue 7, pp 1429–1434, which is incorporated herein by reference. 
     FIG. 23B  illustrates one arrangement of transmission lines  132 ,  134 ,  136 , and  138  on a printed circuit board  120  according to one embodiment of the present invention.  FIG. 23C  illustrates a cross sectional view of the printed circuit board  120  across line D–D′. As shown in  FIGS. 23B and 23C , the first pair of transmission lines  132  and  134  and the second pair of transmission lines  136  and  138  are etched metal lines formed on different layers of the printed circuit board  120 , with the second pair of transmission lines  136  and  138  being directly over the first pair of transmission lines  132  and  134  and separated from the first pair of transmission lines  132  and  134  by a layer of dielectric material  135 . This allows efficient coupling of the input signal from the second pair of transmission lines  136  and  138  into the first pair of transmission lines  132  and  134 . The circuitous routes taken by transmission lines  132 ,  134   136 , and  138  in  FIG. 23B  are just one way to fit the quarter-wavelength long transmission lines into the space allowed on the printed circuit board  120 . The transmission lines can be shaped differently. 
   In one embodiment of the present invention, as shown in  FIG. 24A , antenna  100  is built on the same printed circuit board  120  as the RFID reader employing the antenna for transmitting the interrogation signals and for receiving the responding signals from RFID tags. While antenna  100  only requires one layer of etched metal lines as conductor lines  100   a ,  100   b , and  102 , the RFID reader often uses multiple internal layers of etched metal lines for interconnecting various components of the RFID reader. Thus, when layers of printed circuit board  120  are pressed together, the part of the printed circuit board  120  for accommodating the reader can be significantly thicker than the part of the printed circuit board  120  for accommodating the antenna, as shown in  FIG. 24B . This may cause bubbles to form in the part of the printed circuit board accommodating the antenna, thus affecting the robustness of the antenna. 
   To solve the problem, interlayer metal planes  142  can be placed in the part of the printed circuit board accommodating the antenna, as shown in  FIG. 24C .  FIG. 24D  illustrates a top-down view of the printed circuit board  120 . As shown in  FIG. 24D , interlayer metal planes  142  are placed in the spaces surrounded by the riser  103 , cross member  104  and downward leg  105  of each of conductor lines  100   a  and  100   b , and in the space between conductor lines  100   a  and  100   b , while keeping a minimum distance of about 2 mm from conductor lines  100   a  and  100   b . Interlayer metal planes  142  help to bring even thickness of the printed circuit board  120  and to make the part of the printed circuit board  120  supporting antenna  100  more robust without drastically affecting the performance of antenna  100 . 
     FIGS. 25 and 26  illustrate simulated and measured effects, respectively, of the inductive shunt or matching stub  102  on the impedance response of antenna  100 . The simulated impedance response with and without the matching stub  102  is shown in  FIG. 25 . The measured impedance response with and without the matching stub  102  is shown in  FIG. 26 . Good qualitative agreement is obtained between the simulation and measurements, although there is some quantitative disagreement in the estimate of resonant frequencies. This discrepancy is probably due to the difficulties of accurately removing the effect of the balun from the measured data to obtain the impedance of the antenna structure. As shown in  FIGS. 25 and 26 , the matching stub  102  acts to transform the relatively low equivalent radiation resistance (in the neighborhood of 5 ohms) of the rest of antenna  100  to a much higher value of around 50 ohms, making it easy to match the antenna to a 50-ohm transmission line used to connect the antenna to the radio. 
     FIG. 27  illustrates measured impedance responses of antenna  100  showing the effect of the matching network formed by the discrete matching components  114 ,  115 , and  116  and the balun  113 . 
   EXAMPLE 2 
   A test antenna was constructed to examine the effects of the dimensions and placement of a plastic enclosure (‘radome’). The dimensions of this antenna are shown in Table 2.  FIG. 28  explains the nomenclature describing the radome configuration. 
   
     
       
             
             
             
           
         
             
               TABLE 2 
             
             
                 
             
             
               Parameter 
               Value 
               Units 
             
             
                 
             
           
           
             
               Substrate 
               FR4 
               (NA) 
             
             
               Substrate thickness 
               710 
               microns 
             
             
               L 
               31 
               mm 
             
             
               H 
               11 
               mm 
             
             
               L1 
               9 
               mm 
             
             
               L2 
               32 
               mm 
             
             
               g 
               3.6 
               mm 
             
             
               d 
               2 
               mm 
             
             
               s 
               3.5 
               mm 
             
             
               w 
               1.3 
               mm 
             
             
               Bottom Cavity 
               0.7 
               mm 
             
             
               Height 
             
             
               Radome 
               3 
               (NA) 
             
             
               dielectric constant 
             
             
                 
             
           
        
       
     
   
   The effects of the radome is examined by simulations using a variety of differing radome configurations. The simulation results for different cases of antenna radome configuration are summarized in Table 3. 
   
     
       
             
             
             
             
             
             
             
           
             
             
             
             
           
         
             
               TABLE 3 
             
             
                 
             
             
                 
                 
                 
                 
                 
               Resonant 
               Re(Zin) at 
             
             
                 
               Top 
               Bottom 
               Top 
               Side 
               Frequency 
               Resonant 
             
             
                 
               Cavity Height 
               Thickness 
               Thickness 
               Thickness 
               [MHz] 
               Frequency 
             
             
               Cases 
               [mm] 
               [mm] 
               [mm] 
               [mm] 
               Im(Zin = 0 Ω) 
               [Ω] 
             
             
                 
             
           
           
             
               1 
               6.6 
               1.30 
               1.30 
               1.30 
               948 
               60 
             
             
               2 
               6.6 
               0.65 
               0.65 
               0.65 
               974 
               67 
             
             
               3 
               6.6 
               1.30 
               0.65 
               0.65 
               961 
               61 
             
             
               4 
               6.6 
               0.65 
               1.30 
               1.30 
               962 
               65 
             
             
               5 
               6.6 
               0.65 
               0.65 
               1.30 
               965 
               66 
             
             
               6 
               2.3 
               1.30 
               1.30 
               1.30 
               943 
               55 
             
             
               7 
               2.3 
               0.65 
               0.65 
               0.65 
               972 
               63 
             
           
        
         
             
               8 
               No Radome 
               1011  
               76 
             
             
                 
             
           
        
       
     
   
   A regression fit to the simulations for the resonant frequency is given in Table 4, and a regression fit to the simulations for the real input impedance in Table 5. In each case the factors have been normalized so that their values vary from −1 to +1 and that the effects of each variable can be directly compared. The case with no radome is included as having zero wall thickness and median cavity height. Here the standard error is the estimated error in the coefficient value, and the t-ratio is the ratio of the coefficient to the error estimate. Ratios between −1 and 1 indicate that the coefficient in question is not significant; and ratios greater than 3 or smaller than −3 provide good confidence that the coefficient value is meaningful. 
   
     
       
             
           
             
             
             
             
             
           
             
             
             
             
             
           
         
             
               TABLE 4 
             
           
           
             
                 
             
             
               Resonant frequency 
             
           
        
         
             
                 
                 
                 
               standard 
                 
             
             
                 
               Normalized variable 
               Coefficient 
               error 
               t-ratio 
             
             
                 
                 
             
           
        
         
             
                 
               Constant 
               960 
               1.43 
               673 
             
             
                 
               Top cavity height 
               1.64 
               1.46 
               1.1 
             
             
                 
               Bottom thickness 
               −8.26 
               1.30 
               −6.4 
             
             
                 
               Top thickness 
               −1.44 
               2.24 
               −0.6 
             
             
                 
               Side thickness 
               −6.32 
               1.92 
               −3.3 
             
             
                 
                 
             
           
        
       
     
   
   R 2 =98.9% R 2 (adjusted)=97.3% 
   s=3.4 with 8−5=3 degrees of freedom 
   It is clear that the largest effects of the radome geometry on the resonant frequency result from changes in the bottom and sidewall thicknesses. The real part of the input impedance is mostly affected by the bottom thickness of the radome, with a more modest effect from the top cavity height. 
                                                               TABLE 5                   Real part of input imdedance at resonance                        standard               Normalized variable   Coefficient   error   t-ratio                            Constant   61.3   0.52   117           Top cavity height   2.04   0.54   3.8           Bottom thickness   −3.28   0.48   −6.9           Top thickness   −0.31   0.82   −0.4           Side thickness   −0.97   0.70   −1.4                        
R 2 =98.3% R 2 (adjusted)=95.9%
 
s=1.2 with 8−5=3 degrees of freedom
 
   While the invention has been described with respect to a specific implementation at a specific frequency, it will be appreciated that the inventive principles can be applied by persons of ordinary skill to a wide variety of related applications in which compact, broadside-radiating antennas with good tolerance of ambient variation need to be employed.