Abstract:
Plural-conversion radio receivers for receiving DTV signals, in accordance with the Advanced Television Systems Committee (ATSC) standard, or analog TV, in accordance with the National Television Sub-Committee (NTSC) standard, utilize a first intermediate-frequency band spanning 917-923 MHz and a second intermediate-frequency band spanning 35.5-41.5 MHz. A local oscillator generates local oscillations at 958.5 MHz for mixing with signal in the first intermediate-frequency band to generate signal in the second intermediate-frequency band. These local oscillations do not interfere with the aeronautical navigation band or with channel 81 television broadcasting. The second harmonic of sound carrier in the second intermediate-frequency band falls below the 88-108 MHz FM broadcast band.

Description:
This is a complete application filed under 35 U.S.C. 111(a) claiming, pursuant to 35 U.S.C. 119(e)(1), benefit of the filing date of provisional application serial No. 60/034,611 filed Jan. 7, 1997, pursuant to 35 U.S.C. 111(b). 
    
    
     The present invention relates to the radio receiver portions of television (TV) signal receivers for receiving terrestrial through-the-air television broadcasting in the United States of A America whether the received signals be digital television signals, in accordance with the Advanced Television Systems Committee (ATSC) standard, or analog television signals, in accordance with the National Television System Committee (NTSC) standard. 
     BACKGROUND OF THE INVENTION 
     The first detector in a television signal receiver converts radio-frequency (RF) signals in a selected one of the television broadcast channels, which channels occupy various 6-MHz-wide portions of the electromagnetic wave frequency spectrum, to intermediate-frequency (IF) signals in one particular 6-MHz-wide portion of that spectrum. This first conversion-is typically carried out by superheterodyning the RF signals, which is to say mixing the RF signals with local oscillations from an oscillator oscillating at a frequency substantially higher than the frequencies in the television channel of highest frequency. The first detector is used to convert a selected RF signal to IF signal in order that up to 60 dB or more amplification can be done in that particular 6-MHz-wide portion of that spectrum using intermediate-frequency amplifiers with fixed, rather than variable, tuning. Amplification of the received signals is necessary to raise them to power levels required for further signal detection operations, such as video detection and sound detection in the case of analog TV signals, and such as symbol decoding in the case of digital TV signals. The first detector usually includes variable tuning elements in the form of preselection filter circuitry for the RF signals to select among the various 6-MHz-wide television channels and in the further form of elements for determining the frequency of the local oscillations used for superheterodyning the RF signals. In. TV receivers of more recent design the local oscillator signals are often generated using a frequency synthesizer in which the local oscillator signals are generated with frequency regulated in adjustable ratio with the fixed frequency of a standard oscillator. 
     Television signal receivers for receiving digital television (DTV) signals that have been described in the prior art use plural-conversion radio receivers wherein DTV signal in a selected one of the ultra-high-frequency (UHF) channels is first up-converted in frequency to first intermediate-frequency signal in a first intermediate-frequency band centered at 920 MHz for amplification in a first intermediate-frequency amplifier. The resulting amplified first intermediate-frequency signal is then down-converted in frequency by mixing it with 876 MHz local oscillations, resulting in a second intermediate-frequency signal. This second intermediate-frequency signal, in a second intermediate-frequency band centered at 44 MHz, is then amplified in a second intermediate-frequency amplifier. The response of the second intermediate-frequency amplifier is then synchrodyned to baseband in DTV signal receivers developed by the Grand Alliance. 
     Radio receivers for receiving DTV signals, in which receivers the final intermediate-frequency signal is somewhere in the 1-8 MHz frequency range, are described by C. B. Patel and the inventor in U.S. Pat. No. 5,479,449 issued Dec. 26, 1995, entitled “DIGITAL VSB DETECTOR WITH BANDPASS PHASE TRACKER, AS FOR INCLUSION IN AN HDTV RECEIVER”, and included herein by reference. The radio receivers specifically described in U.S. Pat. No. 5,479,449 are of triple-conversion type using a 920 MHz analog IF amplifier for first detector response, the first detector being an up-converter, and using a 44 MHz analog IF amplifier for second detector response, the second detector being a down-converter. A third detector is a further down-converter, generating a 1-8 MHz final IF signal as third detector response. This final IF signal is not amplified, but is digitized by an analog-to-digital converter for use in digital circuitry for synchronizing to baseband. The resulting digital baseband signal is equalized and then data-sliced in a symbol decoder. The first intermediate-frequency amplifier in one of the DTV signal receivers described in U.S. Pat. No. 5,479,449 uses a surface-acoustic wave (SAW) filter for establishing the bandwidth of the 920 MHz IF amplifier. 
     The DTV signal receiver developed by the Grand Alliance is of double-conversion type using a 920 MHz analog IF amplifier for first detector response, the first detector being an up-converter, and using a 44 MHz analog IF amplifier for second detector response, the second detector being a down-converter. The amplified response of the 44 MHz analog IF amplifier is the final IF signal, which is synchrodyned to baseband in the analog regime. The resulting analog baseband response is then digitized by an analog-to-digital converter prior to being equalized and then data-sliced in a symbol decoder. The first intermediate-frequency amplifier in the DTV signal receiver developed by the Grand Alliance uses ceramic resonators for establishing the bandwidth of the 920 MHz IF amplifiers. 
     For a period of years while DTV broadcasting is becoming established, it is planned that the broadcasting of analog TV signals will continue in the United States in accordance with the NTSC standard using the same UHF channels as DTV signals as well as other channels in the VHF and UHF bands. While analog and digital TV signals occupy the same television channels, the requirements of radio receivers for the two types of TV signal are not particularly compatible. This is pointed out by the inventor in his U.S. patent application Ser. No. 08/825,711 filed Mar. 19, 1997, entitled “RADIO RECEIVER DETECTING DIGITAL AND ANALOG TELEVISION RADIO-FREQUENCY SIGNALS WITH SINGLE FIRST DETECTOR”, and incorporated herein by reference. In that application the inventor points out that the cost of a first detector is substantial enough that it is undesirable to use separate first detectors for analog TV signals and for digital TV signals in radio receivers designed to receive both types of signal, whether those radio receivers are included in a TV set complete with viewscreen or in a digital recording apparatus, such as one using magnetic tape as a recording medium. The use of a single first detector for both analog TV signals and digital TV signals is also desirable in that it allows more-compact radio receiver design and at the same time avoids any problems of unwanted radiation from the output of one of separate respective first detectors for analog TV signals and for digital TV signals to the other first detector. The different radio receiver requirements for analog TV signals and for digital TV signals are accommodated by using different intermediate-frequency amplification for analog TV signals and for digital TV signals. 
     Using different intermediate-frequency amplification for analog TV signals and for digital TV signals allows for the different radio receiver passbands required for each type of TV signal. In an analog TV signal the video carrier is located at a frequency 1.25 MHz above the lower limit frequency of the TV channel, and the vestigial sideband exhibits no gain reduction vis-a-vis the full sideband until modulating frequencies exceed 750 kHz. Accordingly, the radio receiver for an analog TV signal customarily exhibits a linear roll-off of the overall intermediate-frequency response supplied to the video detector, which roll-off is down 6 dB at the video carrier frequency and provides for an overall flat baseband video response up to 4.2 MHz or so. In a DTV signal, the data is located at a frequency only 310 kHz above the lower limit frequency of the TV channel; and roll-off down 6 dB at the data carrier frequency is provided at the transmitter, rather than at the receiver. The overall intermediate-frequency response is essentially flat over a frequency band 6 MHz-wide between 1-dB-down limit frequencies in Grand Alliance receiver designs published by Zenith Radio Corporation. (However, in U.S. Pat. No. 5,786,870 issued Jul. 28, 1998, entitled “NTSC VIDEO SIGNAL RECEIVERS WITH REDUCED SENSITIVITY TO INTERFERENCE FROM CO-CHANNEL DIGITAL TELEVISION SIGNALS”, the inventor describes an analog TV receiver with IF amplifiers providing flat frequency response extending through the carrier, so complex synchronous detection can be employed that better suppresses co-channel DTV signal interference.) 
     A radio receiver for an analog TV signal customarily uses a trap filter for removing frequency-modulated sound carrier from the IF signal supplied the video detector. This is necessary to suppress a 920 kHz beat between the FM sound carrier and the amplitude-modulated chrominance subcarrier, which beat causes unwanted variation in the luminance component of the composite video signal recovered by the video detector. This luminance variation is obtrusively apparent when viewing images reproduced on a television viewscreen. Sound trap filters have not been used in prior-art DTV receiver designs, though co-channel interfering NTSC signals are a known problem during HDTV reception. The avoidance of trap filtering in the IF amplifiers of a DTV signal receiver makes it easier to maintain phase linearity throughout the IF passband. (However, in U.S. patent application Ser. No. 08/826,790 filed Mar. 24, 1997 and entitled “DTV RECEIVER WITH FILTER IN IF CIRCUITRY TO SUPPRESS FM SOUND CARRIER OF NTSC CO-CHANNEL INTERFERING SIGNAL”, which claims priority from a similarly titled provisional application Ser. No. 60/031,358 filed Nov. 20, 1996, the inventor advocates the use of sound trap filters in DTV IF amplifiers to facilitate certain types of comb filters being used to suppress NTSC co-channel interference in the baseband symbol code.) 
     The difference in preferred designs of automatic gain control (AGC) for the radio receiver portions of analog TV signal receivers and of DTV signal receivers is another reason for using different intermediate-frequency amplification for the two types of TV signal. The power in an analog TV signal must be quite high in order that accompanying Johnson or galactic noise is low enough in amplitude as not to cause “snow” (luminance noise) in a black-and-white TV picture or “colored snow”(luminance plus chrominance noise) in a color TV picture. The effective radiated power from an analog TV transmitter is typically tens of kilowatts. The IF amplifiers in an analog TV signal receiver typically provide maximum gain of 60 to 90 dB, which can be reduced responsive to automatic gain control (AGC). Gain reduction of as much as 66 dB is required to handle the gamut of usable signal strengths. When receiving analog TV signals, this gain reduction is preferably obtained using forward AGC in at least the earlier IF amplifier stages. This avoids the problem of internally generated noise in the IF amplifier stages rising vis-a-vis Johnson noise to adversely affect overall noise figure for the radio receiver, which problem is encountered when using reverse AGC. The great concern with loss in noise figure when receiving analog TV signals arises because the human eye is quite sensitive to the presence of random noise accompanying the composite video signal from the video detector. The amplitude of the luminance signal component of the composite video signal directly controls the intensity of light emanating from or reflected from the television display device, and the amplitudes of the chrominance signal component of the composite video signal directly affect the hue and color saturation of that light. 
     In a DTV receiver the radio receiver portion thereof supplies plural-level symbol codes as baseband output signal, and the light emanating from or reflected from the television display device is not directly controlled by the amplitude of such baseband output signal. Small amounts of random noise are strongly rejected by quantizing effects in the data-slicing and trellis decoding associated with symbol decoding. Consequently, the overall noise figure for the radio receiver becomes of concern chiefly when distinguishing between the various levels of the symbol codes becomes a problem. In order best to facilitate distinguishing between the various levels of the symbol codes, linearity of the baseband output signal detected by the radio receiver becomes an important concern, and there is less concern for the overall noise figure for the radio receiver unless long-distance reception of DTV signals is sought for transmissions with power levels in the few hundreds of watts. 
     The AGC of the IF amplifiers in a DTV signal receiver must be such as to avoid non-linearity. Forward AGC tends to introduce non-linearity into the modulation of the IF signal. The resulting distortion is generally tolerable in analog TV signal reception, since larger amplitude modulation properly occurs primarily during synchronizing pulses, and since luminance signal varies in inverse logarithmic relation to scene brightness. Reverse AGC that does not introduce non-linearities into the modulation of the IF signal can be designed for a DTV signal receiver. This can be done using variable-resistance emitter degeneration in a common-emitter transistor amplifier, for example. Or, by way of further example, the collector current of a common-emitter transistor amplifier can be split using common-base transistor amplifiers connected at their emitter electrodes to form a variable-transconductance multiplier. The loss in noise figure with reduction of gain in such reverse AGC arrangements presents little problem as long as overall noise internally generated within the IF amplifier chain of the DTV receiver is smaller than the smallest transitions between digital modulation levels in the final IF amplifier output signal. 
     In prior-art DTV receivers a second conversion, a downconversion from an initial high-intermediate-frequency band to a subsequent low-intermediate-frequency band, is performed using local oscillations at a fixed frequency that is below the initial high-IF intermediate-frequency band. A first intermediate-frequency band that spans 917-923 MHz and a local oscillator that generates oscillations at a frequency of 876 MHz are used in the prior-art DTV receivers. The first IF band is sufficiently higher than the 890 MHz upper limit frequency of the top channel  83  that regeneration of the superheterodyning circuitry used for first detection is not difficult to avoid. However, the 876 MHz local oscillations undesirably fall within channel  81  of the TV broadcast band. 
     The second conversion from the first intermediate-frequency band to the second intermediate-frequency band can be a downconversion using local oscillations at a fixed frequency that is above the first intermediate-frequency band. The reversal of the channel frequency spectrum in the second intermediate-frequency band that obtains in prior-art DTV receivers does not obtain when the local oscillations are at a fixed frequency that is above the first intermediate-frequency band. 
     Local oscillations at 964 MHz can convert from a 917-923 MHz first intermediate-frequency band to a 41-47 MHz second intermediate-frequency band, and such 964 Mlz local oscillations are desirably below the 978-1723 MHz band of local oscillations used in the first detector. The use of 964 MHz local oscillations can give rise to certain interference problems, however. The 964 MHz local oscillations fall within an aeronautical navigation band, so the local oscillator must be carefully shielded. The NTSC audio carrier is translated to 46.75 MHz; and the second harmonic distortion of 46.75 audio carrier occasioned by forward AGC generates an FM carrier at 93.5 MHz, which falls within the FM broadcast band. 
     The use of a low-IF band of 41-47 MHz is convenient at the present time because of the current availability of filters for NTSC signals in this band if one seeks to rely on low-IF-band filtering for establishing the overall frequency selectivity characteristics of the radio receiver for DTV, as is done in the Grand Alliance receivers. The IF band in a single conversion TV receiver is made as high in frequency as practically possible while staying below the tuning range of the preselection filtering in the superheterodyne circuitry used for first detection, in order to reduce the problem that the preselection filtering has with suppressing response to image frequencies that are not at a far remove from the desired frequencies being tuned to. The use of surface-acoustic-wave filters (SAW filters) in the initial high-IF band for establishing the overall frequency selectivity characteristics of the IF amplifier chains eases the problems associated with designing IF amplifiers in a subsequent IF band of reduced frequency. 
     Practically speaking, the initial conversion to frequencies above the band tuned by preselection filtering avoids the need to keep final intermediate frequencies as high as possible to avoid problems with image radio-frequency signals not being sufficiently selected against by the preselection filtering in the superheterodyne circuitry used for first detection. The subsequent conversion can be to lower final intermediate frequencies since images do not fall into the band tuned by preselection filtering. 
     Reducing the second intermediate frequencies sufficiently to translate the frequency-modulated sound carrier to 41.25 MHz sound carrier causes the second harmonic distortion of this carrier occasioned by forward AGC to give rise to an FM carrier at 82.5 MHz, just as in prior-art, which FM carrier falls below the 88-108 MHz FM broadcast band so as not to interfere with any nearby FM broadcast receiver. The local oscillations have to be at 958.5 MHz for mixing with 917.25 MHz sound carrier in the first intermediate-frequency band to generate 41.25 MHz sound carrier in the second intermediate-frequency band. These 958.5 MHz local oscillations fall into a 940-960 MHz band for fixed frequencies that is below the 960-1215 MHz aeronautical navigation band. These 958.5 MHz local oscillations are desirably further below the 978-1723 MHz band of local oscillations used in the first detector. 
     SUMMARY OF THE INVENTION 
     A plural-conversion radio receiver for receiving DTV signals, in accordance with the Advanced Television Systems Committee (ATSC) standard, or analog TV, in accordance with the National Television System Committee (NTSC) standard, embodies a principal aspect of the invention by performing a second conversion from a first intermediate-frequency band to a second intermediate-frequency band using local oscillations at a fixed frequency that is above the first intermediate-frequency band. In radio receivers that are preferred embodiments of the invention, the first intermediate-frequency band spans 917-923 MHz and the second intermediate-frequency band spans 35.5-41.5 MHz. A local oscillator generates local oscillations at 958.5 MHz for mixing with signal in the first intermediate-frequency band to generate signal in the second intermediate-frequency band. These local oscillations do not interfere with the aeronautical navigation band or with channel  81  television broadcasting. The second harmonic of sound carrier in the second intermediate-frequency band, as may be generated in response to intermediate-frequency amplifiers in the application of automatic gain control to the plural-conversion radio receiver, falls below the 88-108 MHz FM broadcast band so as not to interfere with any nearby FM broadcast receiver. The frequency-modulated 41.25 MHz sound carrier in the 35.5-41.5 MHz second intermediate-frequency band corresponds to that in the 41-47 MHz second intermediate-frequency band of prior-art television receivers, but the translated video carrier is 4.5 MHz lower in frequency than 41.25 MHz, rather than 4.5 MHz higher as in prior-art television receivers. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     FIG. 1 is a block schematic diagram of portions of a receiver for analog TV and digital TV signals, which portions of a receiver embody the invention. 
     FIG. 2 is a block schematic diagram of portions of another receiver for analog TV and digital TV signals, which portions of a receiver embody the invention. 
     FIG. 3 is a block schematic diagram of portions of still another receiver for analog TV and digital TV signals, which portions of a receiver embody the invention. 
     FIG. 4 is a block schematic diagram of modifications made of the FIG. 3 receiver portions in another embodiment of the invention in which there are parallel IF amplifiers for the video and audio portions of NTSC analog TV signal. 
     FIG. 5 is a block schematic diagram of modifications made of the FIG. 4 receiver portions in an embodiment of the invention in which the sound component of the NTSC analog TV signal is amplified by 920 MHz and 38.5 MHz IF amplifiers in the so-called “quasi-parallel” arrangement. 
     FIG. 6 is a block schematic diagram of modifications made of the FIG. 5 receiver portions in another embodiment of the invention in which the sound component of the NTSC analog TV signal is amplified by just a 920 MHz IF amplifier in the so-called “quasi-parallel” arrangement. 
     FIG. 7 is a block schematic diagram of the remaining portions of the receiver of any of FIGS. 1,  2 ,  3 ,  4 ,  5  and  6 . 
    
    
     DETAILED DESCRIPTION 
     FIG. 1 shows the radio receiver portions of a television receiver capable of receiving either analog TV signals or digital TV signals. An antenna  1  is a representative source of television signals in the very high frequency (VHF) and ultra high frequency (UHF) bands for application to a first detector  2 . The first detector  2  typically comprises a broadband radio-frequency (RF) amplifier provided with a tracking preselection filter, a first local oscillator with automatic fine tuning, a frequency synthesizer for generating superheterodyning signal of a frequency in selected ratio with the frequency of the first local oscillator, an initial mixer for mixing the selected radio-frequency signal with the superheterodyning signal to generate a first intermediate-frequency signal, and a front-end filter for suppressing the image of the first intermediate-frequency signal in the output signal supplied from the initial mixer. The initial mixer is preferably of a doubly-balanced linear-multiplication type. The first intermediate-frequency signal translates the 6-MHz-wide selected radio-frequency signal so as to be nominally centered at 920 MHz, placing the image frequencies well above 1 GHz so they are easily rejected by a front-end filter with fixed tuning. That is, the first detector  2  can be similar to prior-art first detectors in plural-conversion digital high-definition digital television (HDTV) receivers used by the Grand Alliance during field testing of terrestrial over-the-air HDTV broadcasting in accordance with the ATSC standard. 
     The portion of the FIG. 1 radio receiver specifically for digital TV signals, at the right of the figure, can be similar to those described in U.S. Pat. No. 5,479,449. The first intermediate-frequency signal supplied from the first detector  2  is applied via a buffer amplifier  03  to a surface-acoustic-wave filter  3  having a substantially linear-phase, flat-amplitude response that has −1 dB to −1 dB bandwidth of 6 MHz. The buffer amplifier  03  provides fixed gain to make up the 10-12 dB insertion loss of the SAW filter  3  and drives the SAW filter  3  from a fixed source impedance chosen to avoid unwanted reflections. The response of the SAW filter  3  to this first IF signal is amplified in a wideband intermediate-frequency amplifier  4  that has reverse AGC. The resulting amplified first intermediate-frequency signal from the IF amplifier  4  and local oscillations from a second local oscillator  5  are applied as first and second input signals, respectively, to a first (IF) mixer  6 , which is preferably of a doubly-balanced linear-multiplication type. The second local oscillator  5  is preferably of a type supplying local oscillations at 958.5 MHz frequency, conditioning the mixer  6  to downconvert a 917-923 MHz IF band to a 36.5-42.5 MHz IF band, with pilot carrier for DTV signal at 36.81 MHz, video carrier for NTSC signal at 37.75 MHz, and audio carrier for NTSC signal at 41.25 MHz. The second harmonic of the 41.25 FM sound carrier falls just below the FM broadcast band and does not present an interference problem in a broadcast FM receiver placed near the television receiver. 
     The mixer  6  supplies a second intermediate-frequency signal as its output signal. This second IF signal is applied as input signal to a surface-acoustic-wave filter  7  having a substantially linear-phase, flat-amplitude response over a bandwidth in excess of 6 MHz. The mixer  6  is designed so that its output impedance provides optimal source impedance to the SAW filter  7 , in order to avoid unwanted reflections. The response of the SAW filter  7  is amplified in a wideband intermediate-frequency amplifier  8  that has reverse AGC. The resulting amplified second intermediate-frequency signal from the IF amplifier  8  is synchrodyned to baseband by synchronizing circuitry  9 , and the resulting in-phase (or real) baseband signal is equalized by equalizer circuitry  10 . The resulting equalized in-phase baseband signal is supplied to a symbol decoder  11 . The symbol decoder  11  performs data-slicing operations on the equalized baseband signal to recover data supplied to a trellis decoder in portions of the HDTV receiver not shown in FIG.  1 . The in-phase baseband signal from the syncbrodyning circuitry  9  is supplied to delayed automatic gain-control circuitry  12  that generates the AGC signals controlling the gains of the IF amplifiers  4  and  8 . 
     The automatic gain-control circuitry  12  can take any of a number of known forms. In the early Grand Alliance receivers the AGC circuitry for DTV signals used a matched filter responsive to data segment code groups, and of the response of this matched filter was peak detected to develop a basic AGC signal which was then use to develop delayed AGC for the IF amplifier stages. An AGC that responds to average symbol value can be used, as described by Citta et alii in U.S. Pat. No. 5,565,932 entitled “AGC SYSTEM WITH PILOT USING DIGITAL DATA REFERENCE”. The form for AGC circuitry  12  preferred by the inventor is one that detects the direct component of the baseband signal generated by synchrodyning the pilot carrier to baseband during the reception of DTV signals and that during the reception of analog TV signals falls back on envelope detection of the IF amplifier  8  response to develop AGC. This prevents the IF amplifiers  4  and  8  from operating with excessive gain during the reception of analog TV signals, so that video carrier signals can be extracted from the IF amplifier  8  response. The reader is referred to U.S. Pat. No. 5,636,252 entitled “AUTOMATIC GAIN CONTROL OF RADIO RECEIVER FOR RECEIVING DIGITAL HIGH-DEFINITION TELEVISION SIGNALS”, incorporated herein by reference, which patent issued Jun. 3, 1997 to C. B. Patel and the inventor. 
     U.S. Pat. No. 5,479,449 describes the synchrodyning circuitry  9  as including circuitry for converting the amplified second IF signal to a final IF signal somewhere in a 1 to 8 MHz band, an analog-to-digital converter for digitizing the final IF signal, and digital circuitry for completing the synchrodyne to baseband in the digital regime. Alternatively, as in the receivers used by the Grand Alliance during HDTV field testing, the syncbrodyning circuitry  9  can be operative in the analog regime, with the analog baseband signal being digitized by an analog-to-digital converter for application to the equalizer circuitry  10 . The equalizer circuitry  10  is then cascaded with a phase tracker operative at baseband. 
     The portion of the FIG. 1 radio receiver specifically for analog TV signals, at the left of the figure, includes a surface-acoustic-wave filter  13  having a substantially linear-phase response over a bandwidth in excess of 6 MHz, but exhibiting an amplitude roll-off at higher frequencies that is 6 dB down in gain at 921.75 Mlz, the frequency in the first IF signal that the video carrier is translated to by the first detector  2  This roll-off makes it easier to suppress beats between the video carrier and the sound carrier of the adjacent TV channel. The first IF signal supplied from the first detector  2  is applied to the SAW filter  13  via a buffer amplifier  013  which provides fixed gain to make up the 10-12 dB insertion loss of the SAW filter  13  and drives the SAW filter  13  from a fixed source impedance chosen to avoid unwanted reflections. The response of the SAW filter  13  to the first IF signal is amplified in a wideband intermediate-frequency amplifier  14  that has forward AGC. The resulting amplified first intermediate-frequency signal from the IF amplifier  14  and 958.5 MHz oscillations from a second local oscillator  5  are applied as first and second input signals, respectively, to a second (IF) mixer  15 , which is preferably of a doubly-balanced linear-multiplication type. The mixer  15  supplies a third intermediate-frequency signal as its output signal. This third IF signal is applied as input signal to a surface-acoustic-wave filter  16  having a substantially linear-phase, flat-amplitude response over a bandwidth in excess of 6 MHz. The mixer  15  is designed so that its output impedance provides optimal source impedance to the SAW filter  16 , in order to avoid unwanted reflections. The response of the SAW filter  16  is amplified in a wideband intermediate-frequency amplifier  17  that has reverse AGC. The resulting amplified third intermediate-frequency signal from the IF amplifier  17  is supplied to a surface-acoustic-wave filter  18  that passes frequencies below 41 MHz for application as input signal to a video detector  19 , but traps the FM sound carrier above 41 MHz to prevent 920 kHz beats with chroma subcarrier in the composite video signal detected by the video detector  19 . 
     The composite video signal detected by the video detector  19  is supplied to delayed automatic gain-control circuitry  20  that generates the AGC signals controlling the gains of the IF amplifiers  14  and  17 . The video detector  19  can be either a synchronous detector or an envelope detector. Or, the video detector  19  can comprise a synchronous detector for supplying composite video signal to the luminance and chrominance separation circuitry of the receiver and can further comprise an envelope detector for supplying composite video signal to the AGC circuitry  20  and the sync separation circuitry of the receiver. Usually, a synchronous detector detects with a higher degree of linearity than an envelope detector and better suppresses Johnson noise, justifying the higher cost of synchronous detection as long as an appreciable portion of TV broadcasting is done in accordance with NTSC standards. 
     The amplified third intermediate-frequency signal from the IF amplifier  17  is supplied to a surface-acoustic-wave filter  21  via a buffer amplifier  021  which drives the SAW filter  21  from a fixed source impedance chosen to avoid unwanted reflections. The SAW filter  21  passes frequencies between 41.0 and 41.5 MHz for application as first input signal to a third (IF) mixer  22 , to supply the mixer  22  with FM sound carrier when NTSC analog TV signals are received. The mixer  22  receives, as its second input signal, response from a narrowband bandpass filter  23 . The filter  23  supplies 36.75 MHz video carrier in response to the amplified second IF signal output from the IF amplifier  8 . When NTSC analog TV signals are being received, the output signal from the mixer  22  is a frequency-modulated 4.5 MHz intercarrier; and, when NTSC analog TV signals are not being received, the output signal from the mixer  22  is noise. The output signal from the mixer  22  is amplified in a high-gain sound IF amplifier  24  designed to limit only when frequency-modulated 4.5 MHz intercarrier is present in that signal. The response of the sound IF amplifier  24  is supplied to a frequency discriminator or frequency-modulation detector  25 , which reproduces NTSC composite sound signal. This NTSC composite sound signal is a baseband signal comprising a main channel component that is a left-plus-right signal during stereophonic sound transmissions. During stereophonic sound transmissions the NTSC composite sound signal comprises a stereophonic subcarrier amplitude modulated by a left-minus-right signal. The NTSC composite sound signal may also comprise other subcarriers modulated by subsidiary audio program (SAP) signal(s). 
     One skilled in the art of analog TV receiver design will understand that the SAW filter  18  can be replaced with other types of sound trap filtering, such as the bridged-T by way of example, which other types of sound trap filtering are considered to be equivalents of the SAW filter  18  insofar as the invention is concerned. One skilled in the art of analog TV receiver design will also understand that the SAW filter  18  can be replaced with other types of sound IF selection filtering, such as a double-tuned transformer by way of example, which other types of sound IF selection filtering are considered to be equivalents of the SAW filter  21  insofar as the invention is concerned. One skilled in the art of analog TV receiver design will also appreciate that variants in which the IF amplifier  17 , as well as the IF amplifier  14 , have forward AGC are alternative embodiments of the invention. One skilled in the art of analog TV receiver design will also understand that in alternative embodiments of the invention the SAW filter  13  is replaced by wideband coupling of the first detector  4  output port to the IF amplifier  14  input port, with the low-frequency-end roll-off of the third IF signal being introduced by the SAW filter  16 . If the SAW filter  3  does not have sound trap filtering therein, in other embodiments of the invention the SAW an filter  3  response can be applied to the IF amplifier  14  as its input signal and the SAW filter  13  dispensed with. 
     The fact that frequency-modulated 4.5 MHz intercarrier signal is present in output signal from the mixer  22  only when NTSC analog TV signals are being received ether intentionally or because of strong co-channel interference during DTV signal reception is exploited in the FIG. 1 circuitry to develop a NTSC/ATSC CONTROL signal. Amplified mixer  22  output signal is supplied from the sound IF amplifier  24  to an intercarrier amplitude detector  26 , which detects the average amplitude of the 4.5 MHz intercarrier. The intercarrier amplitude detector  26  can be a simple envelope detector with a time constant of several NTSC scan lines, for example. The baseband response of the amplitude detector  26  is supplied to a threshold detector  27 , which generates the NTSC/ATSC CONTROL signal as its output signal. The threshold detector  27  provides an indication of probable NTSC signal reception, if the detected intercarrier signal exceeds a threshold value in amplitude, and otherwise provides an indication of probable DTV signal reception free from appreciable co-channel interference. 
     In U.S. patent application Ser. No. 08/826,790 filed Mar. 24, 1997 and entitled “DTV RECEIVER WITH FILTER IN I-F CIRCUITRY TO SUPPRESS FM SOUND CARRIER OF NTSC CO-CHANNEL INTERFERING SIGNAL” the inventor teaches that it is preferable that the SAW filter  3  have a substantially linearphase, flat-amplitude response that suppresses the frequency-modulated NTSC sound carrier and that accordingly has a −1 dB to −1 dB bandwidth of only 5.7 MHz. A preferred form for the symbol decoder  11  is also described in that patent application, which is incorporated herein by reference. With the skirts of the SAW filter  3  response being so critically located within the frequency spectrum, automatic fine tuning (AFT) of the first local oscillator in the first detector  2  becomes practically a necessity. The 958.5 MHz local oscillator  5  is crystal stabilized, so that the amplified second IF signal from the second IF amplifier  8  can be used for AFT during the reception of DTV signals. The fact that the responses of the SAW filters  3  and  7  are both amplitude-flat and phase-linear to the edge of the band at which the pilot carrier of the DTV signal and the video carrier of an NTSC analog TV signal are located makes the amplified second IF signal from the second IF amplifier  8  a suitable signal source for the generation of AFT signals no matter whether the TV signal being currently received is a DTV signal or an analog TV signal. 
     When a DTV signal is received, a narrow bandpass filter  28  selects 35.81 MHz pilot carrier from the amplified second IF signal to an AFT detector  29 . The AFT detector  29  is similar in its general construction to those previously used in analog TV signal receivers, typically comprising a limiter amplifier for the bandpass filter response it receives as input signal, a phase shifter for shifting the pilot carrier 90° when it is at prescribed 35.81 MHz frequency, a multiplier for multiplying the differentially phase-shifted pilot carrier signals together, and a lowpass filter for extracting the AFT signal from the resulting product. 
     When an analog TV signal is received, the narrow bandpass filter  23  selects 36.75 MHz video carrier from the amplified second IF signal to an AFT detector  30 . The AFT detector  30  is similar in its general construction to those previously used in analog TV signal receivers, typically comprising a limiter amplifier for the bandpass filter response it receives as input signal, a phase shifter for shifting the pilot carrier 90° when it is at prescribed 36.75 MHz frequency, a multiplier for multiplying the differentially phase-shifted pilot carrier signals together, and a lowpass filter for extracting the AFT signal from the resulting product. 
     An AFT selector  31  selects the AFT signal from the AFT detector  30  for application to the first local oscillator in the first detector  2  when the NTSC/ATSC control signal supplied from the threshold detector  27  indicates that an analog TV signal is being received. The AFT selector  31  selects the AFT signal from the AFT detector  29  for application to the first local oscillator in the first detector  2  when the NTSC/ATSC control signal does not indicate that an analog TV signal is being received. The operation of the dual-mode AFT is described in greater detail and claimed in U.S. patent application Ser. No. 08/822,736 filed Mar. 24, 1997 and entitled “AUTOMATIC FINE TUNING OF TV RECEIVER FOR RECEIVING BOTH DIGITAL AND ANALOG TV SIGNALS”. 
     FIG. 2 shows portions of a radio receiver for receiving analog TV and digital TV signals, which radio receiver portions differ from those in FIG. 1 in the following regards. The SAW filter  13  with −6 dB roll-off in gain at 921.75 MHz is replaced by in-channel and adjacent-channel sound trap filtering  32 , as can be constructed using inductors and capacitors. The cascade connection of SAW filter  16  and wideband IF amplifier  17  with reverse AGC is replaced by an IF amplifier  33  with a passband centered at about 44 MHz and with automatic gain control. The amplified response of the IF amplifier  33  supplies input signals to the SAW filters  18  and  21 . The video SAW filter  18  with sound carrier rejection, but with flat-amplitude response at the low end of the 38.5 MHz IF band is replaced by a video SAW filter  34  providing −6 dB roll-off in overall IF gain at 36.75 MHz as well as sound carrier rejection above 41 MHz. 
     FIG. 3 shows modifications of the FIG. 2 portions of a radio receiver, which modifications provide partially parallel IF amplifier chains for the NTSC video carrier modulation and for the NTSC audio carrier modulation. The SAW filters  18  and  22  are supplied input signals directly from the second mixer  15 , and the wideband IF amplifier  33  is replaced by two wideband IF amplifiers  34  and  35  similar in their respective construction. The IF amplifier  34  amplifies the response of the SAW filter  18  to NTSC video carrier modulation for application to the video detector  19 , and the IF amplifier  35  amplifies the response of the SAW filter  22  to NTSC audio carrier modulation for application to the third mixer  22 . The IF amplifiers  34  and  35  are subjected to corresponding automatic gain control signal from the AGC circuitry  20  so that their respective gains track each other. 
     FIG. 4 shows modifications of the FIG. 3 portions of a radio receiver, which modifications provide completely parallel IF amplifier chains for the NTSC video carrier modulation and for the NTSC audio carrier modulation. The SAW filters  18  and  22  in the 44 MHz IF band are dispensed with. The −6 dB roll-off of video carrier is introduced by a surface-acoustic-wave filter  36 , which replaces the in-channel and adjacent-channel sound trap filtering  32 . The first IF signal supplied from the first detector  2  is applied to the SAW filter  36  via a buffer amplifier  036  which provides fixed gain to make up the 10-12 dB insertion loss of the SAW filter  36  and drives the SAW filter  36  from a fixed source impedance chosen to avoid unwanted reflections. The SAW filter  36  differs from the SAW filter  13  of FIG. 1 in having an in-channel sound trap suppressing response for frequencies below 917.5 MHz. The SAW filter  36  has a substantially linear-phase response over a bandwidth of 5.5 MHz, exhibiting an amplitude roll-off at lower frequencies that is 6 dB down in gain at 921.75 MHz, the frequency in the first IF signal that the video carrier is translated to by the first detector  2 . 
     Insofar as the NTSC audio carrier modulation is concerned, a surface-acousticwave filter  37  separates the 917-917.5 MHz frequencies in the 917-923 MHz IF band, as may contain FM sound carrier, for application to a wideband IF amplifier  38  similar in its construction to the IF amplifier  14 . The first IF signal supplied from the first detector  2  is applied to the SAW filter  37  via a buffer amplifier  037  which provides fixed gain to make up the insertion loss of the SAW filter  37  and drives the SAW filter  37  from a fixed source impedance chosen to avoid unwanted reflections. The IF amplifiers  14  and  38  are subjected to corresponding forward automatic gain control so that their respective gains track each other. A third mixer  39  superheterodynes the amplified response of the IF amplifier  38  with 958.5 MHz oscillations from the local oscillator  5 , to generate third mixer output signals in a 41.0 to 41.5 MHz band that are applied as input signal to the wideband IF amplifier  35 . The amplified response of the IF amplifier  35  is supplied to the mixer  22  as its first input signal, the mixer  22  being a fourth mixer in the FIG. 4 circuitry. 
     FIG. 5 shows modifications of the FIG. 4 portions of a radio receiver, in which modifications the sound component of the NTSC analog TV signal is amplified by IF amplifiers arrayed in the so-called “quasi-parallel” arrangements The SAW filter  37  for the 917-917.5 MHz band is replaced by a surface-acoustic-wave filter  40  having a double-hump response peaking at the 917.25 MHz NTSC audio carrier and at the 921.75 MHz NTSC video carrier and at least 10 dB down in response between humps. The first intermediate-frequency signal supplied from the first detector  2  is applied to the SAW filter  40  via a buffer amplifier  040  which provides fixed gain to make up the 10-12 dB insertion loss of the SAW filter  40  and drives the SAW filter  40  from a fixed source impedance chosen to avoid unwanted reflections. The fourth mixer  22  is replaced by a non-linear device operative as a sound detector  41 , such as the rectifier in a simple envelope detector, that generates intercarrier input signal for the 4.5 MHz sound IF amplifier  24 . 
     FIG. 6 shows modifications of the FIG. 5 portions of a radio receiver, which modifications in which the response of the SAW filter  40  having a doublehump response is amplified by a wideband IF amplifier  42  with controlled gain. The response of the IF amplifier  42  is supplied to a non-linear device operative as a sound detector  43 , such as the rectifier in a simple envelope detector, that generates intercarrier input signal for the 4.5 Mz sound IF amplifier  24 . 
     There is no need to extract video carrier from the IF amplifier  8  response during the reception of analog TV signals when the modifications of FIG. 5 or of FIG. 6 are used, so there is no longer a particular necessity to keep the IF amplifiers  4  and  8  from operating with excessive gain when DTV signals are not being currently received. This permits the AGC circuitry  12  to be simplified, since there is no longer a need to fall back on envelope detection of the IF amplifier  8  response to develop AGC when DTV signals are not being currently received. 
     FIG. 7 shows the remaining portions of a television set employing radio receiver portions as shown in any of FIGS. 1,  2 ,  3 ,  4 ,  5  and  6 . In the DTV portion of the receiver, the data recovered in serial-bit form by the symbol decoder  11  are supplied to a data interleaver  43 , which supplies interleaved data in parallel-bit form to trellis decoder circuitry  44 . The trellis decoder circuitry  44  supplies output signal in parallel-bit form to a data de-interleaver  45 , and the output signal from the data de-interleaver  45  are parsed into bytes by parsing circuitry  46  for application to decoder circuitry  47  for decoding Reed-Solomon forward error-correction coding. The output signal from the Reed-Solomon decoder circuitry  47  is supplied to a data de-randomizer  48  which supplies packets of data to a packet sorter  49 . The packet sorter  49  selects packets of video data to an MPEG-2 decoder  50 , which supplies a digital luminance (Y) signal and digital chrominance (U and V) signals to video source chooser circuitry  51  in delayed response to those packets of video data. Further, the packet sorter  49  selects packets of audio data to a digital audio decoder  52 , which generates digital stereophonic audio signals supplied to digital-to-audio converters (DACs)  53  and  54 . The DACs  53  and  54  convert the digital stereophonic audio signals to analog stereophonic audio signals supplied to audio source chooser circuitry  55 . The output signal from the equalizer  10  is supplied to ATSC sync separation circuitry  56  for detecting code groups specifying the beginnings of data fields and of data segments in the datastream. 
     The signals the ATSC sync separation circuitry  56  produces at the beginnings of the data fields are supplied to a controller  57  via a connection  58  and the signals the ATSC sync separation circuitry  56  produces at the beginnings of the data segments are supplied to the controller  57  via a connection  59 . The controller  57  supplies a signal applied via a connection  60  to the video source chooser circuitry  51  for controlling its selection of video source and applied via a connection  61  to the audio source chooser circuitry  55  for controlling its selection of audio source. When DTV signal is being received, the controller  57  conditions the video source chooser circuitry  51  to select the digital luminance (Y) signal and digital chrominance (U and V) signals supplied by the MPEG-2 decoder  50  for application to a display buffer memory  62 , the writing from which memory  58  is controlled by the controller  57  via a control link  63 , and reading into which memory  58  is controlled by the controller  57  via a control link  64 . When DTV signal is being received, the controller  57  conditions the audio source chooser circuitry  55  to select the analog stereophonic audio signals supplied by the DACs  53  and  54  for amplification by the audio amplifiers  65  and  66 , which supply their respective amplifier responses to a left loudspeaker  67  and a right loudspeaker  68 . The controller  57  knows DTV signal is being received when the ATSC sync separation circuitry  56  detects the beginnings of the data fields in a DTV signal. Alternatively, circuitry for detecting the sustained presence of pilot carrier in a DTV signal being currently received can be used to inform the controller  57  of DTV signal reception. 
     The display buffer memory  62  is read from to supply raster-scanned digital luminance (Y) signal to a digital-to-analog converter  69  and digital chrominance (U and V) signals to digital-to-analog converters  70  and  71 . The resulting analog luminance (Y) signal from the digital-to-analog converter  69  and analog chrominance (U and V) signals from the digital-to-analog converters  70  and  71  are supplied to color matrixing circuitry  72  of analog type to generate red, green and blue analog color signals amplified by amplifiers  73 ,  74  and  75 , respectively. The amplified red, green and blue color signals are supplied to a display device  76 , the raster scanning of which is controlled by the controller  57 . In TV receivers alternative to those diagrammed in FIG. 7, the raster-scanned digital luminance (Y) signal and digital chrominance (U and V) signals read from the display buffer memory  62  can be supplied to color matrixing circuitry of digital type to generate red, green and blue digital color signals that are then converted to red, green and blue analog color signals to be amplified by the amplifiers  73 ,  74  and  75 , respectively. 
     In the analog TV portion of the receiver, the video detector  19  supplies composite video signal to NTSC sync separation circuitry  77 , which supplies horizontal and vertical synchronization signals to the controller  57  via connections  78  and  79 , respectively. The video detector  19  also supplies composite video signal to luminance/chrominance separation circuitry  80 , which circuitry  80  separates an analog chrominance subcarrier signal for application to color circuitry  81  and separates an analog baseband luminance signal for application to an analog-to-digital converter  82 . The color circuitry  81  receives a burst gate signal from the NTSC sync separation circuitry  71  via a connection  83  and responds to its input signals to supply an analog color-difference (U) signal to an analog-to-digital converter  84  and to supply another analog color-difference (V) signal to an analog-to-digital converter  85 . The digitized luminance signal is supplied by the ADC  82  to a scan line doubler  86 , which converts the  525  scan lines of NTSC luminance to a luminance signal having 1050 scan lines. The scan line doubler  86 , the ADC  75  and the ADC  76  supply digitized Y, U and V signals to the video source chooser circuitry  51 , for selection to the display buffer memory  62  when the controller  57  determines that an NTSC signal is currently being received and that no DTV signal is currently being received. The FM detector  25  supplies composite audio signal to a stereophonic decoder  87 , which responds to composite audio signal for supplying stereophonic signals to the audio source chooser circuitry  55 . The controller  57  receives the NTSC/ATSC CONTROL signal from the threshold detector  27  and uses it to determine whether or not an NTSC signal is currently being received. When the controller  57  determines that an NTSC signal is currently being received and that no DTV signal is currently being received, the controller  57  conditions the audio source chooser circuitry  55  to supply the audio amplifiers  65  and  66  stereophonic signals responsive to those from the stereophonic decoder  87 . 
     FIG. 7 shows a video recorder  88  being included in combination with the TV set, forming what is known in the industry as a “combo”. FIG. 7 shows the analog stereophonic audio signals from the audio source chooser circuitry  55  and the analog baseband luminance and baseband chrominance signals from the DACs  69 ,  70  and  71  being supplied to the video recorder  88 , there to be digitized again by the video recorder  88  if it is a digital rather than an analog video recorder. The arrangement of video recorder  88  permits recording of NTSC signals as well as DTV signals, which is not possible with a digital video tape recorder arranged to record packets of digital television information before decoding by the MPEG-2 video decoder  50  and the digital audio decoder  52 . There are, of course, embodiments of the invention in which the elements  65 - 68  and  72 - 76  are dispensed with. 
     The embodiments of the invention that have been thusfar described do not use automatic gain control of the RF amplifier in the first detector  2 . The RF amplifier is reasonably broadband and has a gain of only 10 dB or so; gain control would be desired primarily in environments where input signal strength is exceptionally great, such as in a reception location very near a broadcast transmitter. Where AGC of the RF amplifier is undertaken, it is suggested that both forward-AGC and reverse-AGC arrangements be provided for the RF amplifier. When the NTSC/ATSC CONTROL signal from the threshold detector  27  indicates an NTSC signal is currently being received, the reverse AGC mechanism should provide no gain reduction, and the forward AGC should be relied on to control the RF amplifier gain. When the NTSC/ATSC CONTROL signal from the threshold detector  27  indicates an NTSC signal is not currently being received, the forward AGC mechanism should provide no gain reduction, and the reverse AGC should be relied on to control the RF amplifier gain. 
     DTV receivers that receive QAM DTV signals used in cablecasting as well as VSB DTV signals used for terrestrial through-the-air broadcasting are described by C. B. Patel and the inventor in U.S. Pat. No. 5,506,636 issued Apr. 9, 1996, entitled “HDTV SIGNAL RECEIVER WITH IMAGINARY-SAMPLE-PRESENCE DETECTOR FOR QAM/VSB MODE SELECTION”. Where the SAW filter  3  is of a type having a substantially linear-phase, flat-amplitude response that has −1 dB to −1 dB bandwidth of 6 MHz the response of the IF amplifier  8  can be used to supply input signals to circuitry for synchrodyning QAM DTV signals to baseband, as well as to supply input signals to circuitry  9  for synchrodyning VSB DTV signals to baseband. Where the SAW filter  3  is of a type having a narrower bandwidth, so that FM NTSC sound carrier is trapped, the QAM DTV signals will require a parallel IF amplifier chain of their own and a SAW filter of the type having a substantially linear-phase, flat-amplitude response that has −1 dB to −1 dB bandwidth of 6 MHz for selecting the first detector  2  response as input signal to that separate IF amplifier chain. That separate IF amplifier chain can include a mixer that heterodynes an amplified high-IF-band signal with oscillations from the local oscillator  5  to generate a low-IF-band signal. 
     In the claims which follow, the word “said” is used whenever reference is made to an antecedent, and the word “the” is used for grammatical purposes other than to refer back to an antecedent.