Abstract:
When cables run side by side over a longer distance, an alternating-current signal used for cable location can couple or ‘bleedover’ to neighboring cables. The coupled current flowing in the neighboring cable creates field distortion and makes position determination of the targeted cable difficult. The resulting magnetic field of both (or more) cables has a non-circular shape and is commonly known as a distorted field. Established methods for finding the position of the cable under investigation lead to inaccuracies or even wrong locates. The method described herein eliminates the field distortion that is due to the coupled cables by demodulating a phase reference signal placed on the cable by a transmitter into two signal strength constituents. The inphase signal represents the field strength of the targeted conductor and is substantially free of field distortion. The other quadrature signal contains the component of the field associated with distortion.

Description:
BACKGROUND 
   1. Field of the Invention 
   The present invention relates to detection of electromagnetic signals from targeted hidden conductors and, in particular, to the decoupling of interference signals in detected signals from the targeted hidden conductors that result from bleedover of the transmitted signal to adjacent non-targeted conductors. 
   2. Discussion of Related Art 
   Underground pipe and cable locators (sometimes termed line locators) have existed for many years and are well known. Line locator systems typically include a mobile receiver and a transmitter. The transmitter is coupled to a target conductor, either by direct electrical connection or through induction, to provide a current signal on the target conductor. The receiver detects and processes signals resulting from the electromagnetic field generated at the target conductor as a result of the current signal, which can be a continuous wave sinusoidal signal provided to the target conductor by the transmitter. 
   The transmitter is often physically separated from the receiver, with a typical separation distance of several meters or in some cases up to many kilometers. The transmitter couples the current signal, whose frequency can be user chosen from a selectable set of frequencies, to the target conductor. The frequency of the current signal applied to the target conductor can be referred to as the active locate frequency. The target conductor then generates an electromagnetic field at the active locate frequency in response to the current signal. 
   Different location methodologies and underground environments can call for different active frequencies. The typical range of active locate frequencies can be from several Hertz (for location of the target conductor over separation distances between the transmitter and receiver of many kilometers) to 100 kHz or more. Significant radio frequency interference on the electromagentic field detected by the receiver can be present in the environment over this range. Therefore, receivers of line location systems have often included highly tuned filters to preclude interference from outside sources from affecting the measurement of signals at the desired active locate frequency from the target conductor. These filters can be tuned to receive signals resulting from electromagnetic fields at each of the selectable active locate frequencies and reject signals resulting from electromagnetic fields at frequencies other than the active locate frequencies. 
   In line location systems, the signal strength parameter determined from detection of the electromagnetic field provides basis for derived quantities of the current signal (i.e., the line current in the targeted conductor), position of the line locator receiver relative to the center of the conductor, depth of the conductor from the line locator receiver, and can also be used as the input to a peak or null indicator (depending on the orientation of the magnetic field to which that the detector is sensitive). All line location systems measure signal strength on one or more measurement channels. 
   Often in a crowded underground utility environment of metallic pipes and cables, coupling of signals at the active locating frequency from the target conductor to other adjacent underground conductors can occur. These conductors (lines) are not intended to be tracked by the line location system, but coupling of currents from the target conductor to those neighboring conductors through various means (resistive, inductive, or capacitive), termed “bleedover,” can lead a line locator astray such that the operator of the line location system ceases tracking the targeted conductor (e.g., pipe or cable of interest) and instead begins following an adjacent line. 
   In conventional receivers, it is nearly impossible to determine whether the receiver is tracking the targeted conductor or whether the receiver is erroneously tracking a neighboring conductor. In complicated underground conductor topologies, the effect of interference from electromagnetic fields resulting from bleedover currents in neighboring conductors can result in significant assymetrical electromagnetic fields, which is termed field distortion. Further, conventional systems that attempt to distinguish between the targeted conductor and neighboring conductors typically rely on wireless or wired transmission of phase information from the transmitter, which may be located at such a distance from the receiver of the line locator that receiving such information is impractical. 
   Therefore, there is a need for line location systems capable of accurately determining the signal strength parameter from the targeted conductor exclusive of neighboring conductors that may provide signals that are a result of inductive or capacitive coupling, using a signal generation and processing method that utilizes only the targeted conductor (pipe or cable) as the transmission medium without relying on a separate communication channel for the transmitter and receiver to share phase information. 
   SUMMARY 
   In accordance with the present invention, a line locator and line locator system is presented that can distinguish between signals received from a targeted conductor and signals received as a result of bleedover to neighboring conductors. A receiver for a line locator according to some embodiments of the invention includes a first digital phase-locked loop coupled to receive an input signal and lock to a carrier frequency of the input signal; a second digital phase-locked loop coupled to receive an FM signal from the first digital phase-locked loop and lock to a modulation frequency of the input signal; and a quadrature demodulator coupled to receive the input signal and a frequency signal from the second digital phase-locked loop and provide an in-phase signal. 
   In some embodiments of the invention, a line locator system includes a transmitter coupled to provide a current signal to a target conductor, the current signal including at least one signal at a carrier frequency, the signal being modulated at a modulation frequency; and a locator. In some embodiments, the locator includes a detector system that provides at least one signal related to an electromagnetic field present at the locator; at least one receiver coupled to receive the at least one signal, each of the at least one receiver including a first digital phase-locked loop that locks to the carrier frequency of one of the at least one signal, a second digital phase-locked loop coupled to receive an FM signal from the first digital phase-locked loop and lock to the modulation frequency of the one of the at least one signal; and a quadrature demodulator coupled to receive the one of the at least one signal and a frequency signal from the second digital phase-locked loop and provide an in-phase signal; and a display coupled to receive the in-phase signal and provide information to a user. 
   A method of detecting signals associated with a target conductor while rejecting signals associated with bleedover to neighboring conductors according to some embodiments of the present invention includes providing an input signal in response to an electromagnetic field; locking to a carrier frequency of the input signal and providing an FM signal; locking to a modulation frequency in the FM signal signal and providing a frequency signal; and mixing the frequency signal with the input signal to provide an in-phase signal. 
   A method of determining a depth of a target conductor according to some embodiments of the present invention includes providing a current signal on the target conductor, the current signal including a signal at a carrier frequency, the signal being modulated at a modulation frequency; determining a target line signal strength at a plurality of positions, the plurality of positions disposed along a line substantially perpendicular to a line of travel of the target conductor; and determining the depth from the target line signal strength at the plurality of positions. In accordance with some embodiments of the invention, the target line signal strength at each of the plurality of positions can be determined by providing an input signal in response to an electromagnetic field at the position, locking to a carrier frequency of the input signal and providing an FM signal signal, locking to a modulation frequency in the FM signal and providing a frequency signal, mixing the frequency signal with the input signal to provide an in-phase signal, and determining the target line signal strength from the in-phase signal. 
   These and other embodiments are further discussed below with reference to the following figures. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1A  illustrates utilization of a line locator system. 
       FIG. 1B  illustrates an example of electrical circuits that can be created during line location in a particular conductor topology. 
       FIG. 2  shows a simplified equivalent circuit to those shown in  FIG. 1B . 
       FIGS. 3A and 3B  show schematically the equivalent electrical circuit of the situation shown in  FIG. 2  and illustrates the current signals. 
       FIG. 4  illustrates in a block diagram of a receiver according to the present invention. 
       FIG. 5  illustrates a digital phase locked loop that can be utilized in embodiments of a receiver according to the present invention. 
       FIG. 6  illustrates another embodiment of receiver according to the present invention. 
       FIG. 7  is a cross section of an underground locate scenario with only one pipe or cable (directed into and out of the paper). 
       FIGS. 8A and 8B  illustrate determination of line location and depth with two buried conductors separated by an unknown distance. 
   

   In the figures, elements having the same designation have the same or similar functions. Elements in the figures are not drawn to scale. 
   DETAILED DESCRIPTION 
     FIG. 1A  illustrates a line location environment with a line locator. The line locator includes locator  110  and transmitter  101 . Locator  110  can include an embodiment of receiver according to the present invention or may be a conventional locator. An example of a mostly digital implementation of a receiver  110  is described in U.S. application Ser. No. 10/622,376, “Method and Apparatus for Digital Detection of Electromagnetic Signal Strength and Signal Direction in Metallic Pipes and Cables,” by James W. Waite and Johan D. Overby (the &#39;376 application), which is assigned to the same assignee as is the present invention and herein incorporated by reference in its entirety. 
   Locator  110  is operated by operator  116  in order to locate target conductor  104 . Target conductor  104  is directly coupled to transmitter  101 . In many systems, transmitter  101  couples a current signal onto conductor  104 . As discussed above, the current signal has a frequency at the active locate frequency, which can be one of a set of standardized frequencies. As shown in  FIG. 1A , target conductor  104  can be below ground. Transmitter  101  can be coupled to line  104  directly at an above ground junction box or by digging to line  104  in a known location. Transmitter  101  can also be electrically coupled to line  104  inductively. 
   Receiver  110  includes detectors  112  and a display  114 . Detectors  112  can be any detectors for determining an electromagnetic field. For example, detectors  112  can include coils which provide electrical signals in the presence of time varying magnetic fields. Electrical signals can be processed in receiver  110  and the results of the processing can be communicated to operator  116  on display  114 . 
   Detectors  112 , then, can detect magnetic field  118  generated by the current signal in target conductor  104 . However, a current of the same frequency can be coupled into neighboring line  126  by resistive coupling, capacitive coupling, or inductive coupling. Neighboring line  126 , then, can generate magnetic field  120 . The signal provided to locator  110  by detectors  112 , then, will reflect the contributions from both magnetic field  118  from target conductor  104  and magnetic field  120  from neighboring conductor  126 . Calculations of the depth of target line  104  or the current in target line  104 , then, may be inaccurate. 
   Removing interference from neighboring conductor  126  due to resistive coupling is discussed in U.S. Pat. No. 6,411,073, “Method and Device for Locating a Metal Line”, by Volker Fraedrich and Gerhard Fischer (the &#39;073 patent), which is herein incorporated by reference in its entirety. The &#39;073 patent discloses a line location system with an FM modulation method generated by transmitter  101  (called “Signal Select”) that allows transmitter  101  and receiver  110  to share a common phase reference. Subsequently, the &#39;376 application describes a robust method of measurement at the receiver of the signal direction parameter, allowing the operator to detect situations when ground-return path signals are present on adjacent conductors. In those cases, receiver  110  presents a negative signal direction indication to the user, indicating that the measured field strength is due to a resistive bleedover from a conductor carrying a ground return current rather than being due to magnetic field  118  of target conductor  104 . 
   It is often the case that resistive coupling occurs between adjacent lines. If resistive coupling occurs, the induced voltage signal in the adjacent line is reversed (180° out of phase) from the current signal present in the target conductor. This is because the current that has propagated to the adjacent line is seeking an easier (i.e., lower impedance) return path to the same ground stake at the transmitter of the line locating system. By convention, the outgoing signal from transmitter  101  is taken as the positive direction, and the incoming signal to transmitter  101  is taken as the negative direction. By monitoring the signal direction in addition to signal strength, one can detect a likely resistive coupling situation through a positive-to-negative direction change. 
   In the case where bleedover from line  104  to line  126  is inductive or capacitive, the problem is somewhat different. Inductive and capacitive bleedover can occur in long locates (i.e., where receiver  110  is well separated from transmitter  101 ) where target conductor  104  is of good quality and the resistance between ground and the cable sheath of target conductor  104  is high or, alternatively, where the active locate frequency is high (lowering the impedance threshold wherein capacitive coupling becomes a problem). In capacitive or inductive bleedover the signal on the neighboring conductor (e.g., conductor  126 ) undergoes a phase shift that corresponds to the transfer function of the coupling. In situations where the coupling is predominantly inductive or capacitive, the signal is 90° out-of-phase from the current signal that is transmitted by transmitter  101 . 
   Sometimes the bleedover signal (the signal generated from detection of the electromagnetic field from the neighboring conductor) is significantly stronger than the signal generated in receiver  110  by detection of the electromagnetic field from target conductor  104  and completely dominates the measured field strength detected at receiver  110 . A particularly problematic measurement situation occurs when the neighboring cable, for example conductor  126  in  FIG. 1A , which carries a bleedover signal, is shallower than the target conductor. The combined signal at the receiver (e.g., the signal resulting from detection of both magnetic field  118  and magnetic field  120  at receiver  110 ) has the exact same frequency, the active locate frequency, as that of the current signal applied to target conductor  104  by transmitter  101  (it is assumed that the system is Linear Time Invariant (LTI), so the only phase changes observed are due to the bleedover coupling itself). In some cases, the electromagnetic field resulting from the bleedover signal in the shallower neighboring conductor can be stronger than the electromagnetic field generated by target conductor  104 , in which case the measured signal at the receiver will tend toward a 90° phase shift from that imparted to targeted conductor  104  by transmitter  101 . 
   In conventional receivers it is impossible to know or determine the phase of transmitter  101  at receiver  110  because transmitter  101  usually includes a free running oscillator driver. The field strength measured by locator  110 , however, includes the effects of the inphase signals from detection of the electromagnetic field from targeted conductor  104  and the electromagnetic fields from the ground return, which are 180° out-of-phase, due to resistive bleedover, as well as the quadrature signals due to detection of electromagnetic fields from inductively or capacitively coupled lines. In complicated underground utility scenarios with significant coupling, the net magnitude of these signals can result in a non-concentric magnetic field, an effect which is commonly referred to as field distortion. 
   Some prior art line locate systems have circumvented significant field distortion caused by capacitive or inductive bleedover by dropping the active locate frequency to a low enough value that bleedover is avoided. However, for long locates over many kilometers, the active frequency generally needs to be lower than about 10 Hz to avoid significant bleedover. Examples of such systems are described in U.S. Pat. Nos. 5,798,644 and 6,127,827. 
   Other systems detect the presence of field distortion by comparing field strength signals derived from various detector coil configurations. Once detected, several techniques can be used to adjust the locate position of the target conductor for the distortion. A standard peak and null coil configuration is suggested in U.S. Pat. No. 6,215,888, from which the measured field strengths can be compared to models of field strengths stored in a database. More accurate target line positioning is then possible by deducing the most likely underground cable topology that would result in the measured field strengths utilizing the selected model of field strengths. Another method is described in U.S. patent application Ser. No. 10/189,342, by Russell Bigelow (the &#39;342 application), which is assigned to Metrotech Corporation and herein incorporated by reference in its entirety. The &#39;342 application refers to a numerical approach to compare signal strength measurements from three or more coils to those that would result from an undistorted field. Other approaches toward dealing with field distortion is to make use of position measurement methods to map the magnetic field over the surface of the ground and detect non-concentric fields, a good indication of distortion. A line locator that includes an inertial position-tracking device is described in U.S. patent application Ser. No. 10/407,705 “Buried Line Locator with Integral Position Sensing”, by Gordon Pacey and assigned to Metrotech Corporation, which is herein incorporated by reference in its entirety. 
   In such previous systems, the measured field strength parameter, detected from the electromagnetic field at locator  110 , can be a combination of both the radiating targeted conductor as well as other neighboring coupled conductors. When such coupling exists, the degree of distortion present is a primary factor in the overall quality of the target conductor locate. In resistively coupled situations, U.S. Pat. No. 6,411,073 and the &#39;376 application teach that the signal direction parameter is useful to discriminate the targeted conductor from ground return currents present in parallel-coupled conductors. The direction of the signal at the receiver is derived from the Signal Select modulation to the signal imparted at the transmitter thus a common phase reference is available at the receiver. 
   The method described in U.S. Pat. No. 5,194,812 (the &#39;812 patent) accomplishes noise reduction through the wireless (or above ground wired) transmission of the locate signal from the transmitter to the receiver. As long as the wireless link is robust and not itself subject to unknown phase lags, a phase reference is available to the receiver. Synchronous signal detection is achieved at the receiver using an analog mixer followed by a low-pass filter, resulting in a reduction of noise. The &#39;812 patent did not discuss decoupling of distortion caused by inductive or capacitance coupling to other neighboring lines. Also, this method of signal phase reference transmission is prone to signal blockage and has a limited range, which limits its capabilities for long range line location. Time synchronization (and hence phase reference synchronization) between transmitter and receiver via a shared timebase like GPS is also limited by transmission path issues. 
   In accordance with the present invention, a locator  110  is presented that includes a receiver that separates the signal resulting from the detection of the electromagnetic field at receiver  110  into an inphase and a quadrature signal. Thusly, the signals resulting from the current signal applied to the targeted conductor and from resistively coupled bleedover are separated from signals resulting from inductive or capacitive bleedover. Embodiments of the present invention extend the receiver processing associated with Signal Select (as described in the &#39;376 application) demodulation to allow a distinct separation of resistively coupled signals (from the target conductor and ground return path signals) from undesirable (distortion causing) signals caused by inductive or capacitive coupling. 
   The most frequent form of coupling is resistive, of which ground return currents in conductors lying parallel to the target line over the entire length of the locate are the most typical. This is a common problem in city scenarios. The current flowing through a cable returning via other cables (or ground) generates a magnetic field that is sensed with the opposite polarity by a coil antenna of receiver  110 . On long cables, e.g., about 50 miles and longer, it is less likely that a non-targeted conductor lies parallel over the entire course and hence its facility as an easy (i.e., low impedance) ground return path is limited. More likely, close to the start and endpoint, there will be cables going the same direction for a while, but on the majority of the run, the other cables should just be local neighbors of limited co-located length. That implies that any coupling will be primarily inductive. Capacitive coupling, the third possibility, can play a role in dry and low ground-conductivity areas. If the coupling on the majority of the cable run is inductive, then there will be a phase difference of close to 90°, as it is loose coupling via mutual induction. For capacitive coupling, the phase in the coupled circuit will be −90°. 
   In many conventional cable locator receivers, the extracted signal phase has not been used as a measure of the coupling type because a solid phase reference with respect to the transmitter phase has not been achieved. The &#39;376 application notes the development of a nested (dual) digital phase locked loop (DPLL) so that the Signal Select phase reference could be accurately recovered at the receiver. Some embodiments of the present invention described herein utilize a similar approach to recover a stable version of the FM modulation frequency applied by transmitter  101  and, in addition, allow the use of that recovered FM modulation frequency signal as a mixer input to a quadrature demodulator. The quadrature demodulated inphase signal includes all resistively coupled (0° and 180°) signals and the demodulated quadrature signal is representative of the net inductive and capacitive coupling (+/−90°). In addition, the sign of the inphase signal is taken as the signal direction, representing the net direction of the resistively coupled signal, as is described in the &#39;376 application. 
   In some embodiments of the present invention, the processing at locator  110  of the Signal Select modulated locate signal can be further enhanced by averaging and/or low-pass filtering, allowing stable estimates of target line field strength (cleaned of substantially all contribution due to inductively or capacitively coupled lines), total field strength (including that due to all coupled lines), and signal direction. In some of these embodiments, it is not necessary to directly compare phases between two signals of different frequencies to estimate the signal direction. This is an important result in that the accuracy of phase comparison in a digital system is driven in part by the degree of oversampling in the signal processing. Parameter estimation via averaging is more robust than comparison of sample values, and the system can operate reliably for sample rates close to the Nyquist rate (no oversampling). 
   In some embodiments, the field strength values can be presented to user  116  at display  114  in a plot as a function of lateral position x over the targeted line, where x is the coordinate perpendicular to the cable. The areas with large differences between the total field strength and the target conductor field strength indicate the influence of a coupled line or lines. When the coordinate x is reliably obtained via tape measurement, laser range finding, inertial sensing, GPS navigation, or other survey methods, the target line field strength, being effectively cleaned of the influence of coupled lines, can be used to accurately triangulate or vector the coordinates of the underground conductor (both x position and depth). 
   Some embodiments of the invention facilitate the use of non-linear optimization methods to simultaneously estimate the depth, position, and current flowing in targeted conductor  104  from the target line field strength. These estimates are more reliable in the presence of magnetic field distortion due to inductive or capacitive coupling because the target line field strength more closely conforms to the physical model of a concentric field around the targeted conductor  104 . 
   As a further confirmation of the presence of coupled lines and thus distortion, in some embodiments two or more active Signal Select modulated frequencies (one lower and one higher) can be placed on target conductor  104  by transmitter  101 . For coupled lines the impedance drops with increasing frequency, thus the difference between the total field strength and target field strength will be larger at the higher frequency. Some embodiments of the invention described herein can utilize an efficient transmitter algorithm for generating multiple simultaneous Signal Select modulated frequencies on the targeted conductor. To support the comparison of the degree of field distortion at two or more frequencies, the receiver signal processing is expanded to implement multiple parallel Signal Select demodulation algorithms. 
     FIG. 1B  is a depiction of a line locate scenario that involves a forward current from transmitter  101  to splice box, a reverse ground return current from the splice box back to the transmitter ground stake, and an induced current on neighboring line  106 . The conductor topology illustrated in  FIG. 1B  shows a target conductor  104  and a neighboring conductor  106 .  FIG. 1B  also illustrates a direct connection transmitter  101  electronically coupled to target conductor  104 . 
   Line location using direct-connect transmitter  101  utilizes a galvanic connection to the targeted conductor  104  (also referred to as the targeted line) such as is illustrated in  FIG. 1B . Often there is access to targeted conductor  104  through a splice or junction box  102  at the terminus of the line so that the far end of targeted conductor  104  (i.e., the end opposite transmitter  101 ) can be grounded to earth  107 . In the case of telecommunication cables, for example, effective line tracing can be achieved by grounding the metal sheath around the copper or fiber optic cable at the far end, so that a closed loop AC circuit is created as shown by the combination of paths  104  and  107 . 
   Transmitter  101  generates a current signal in target line  101 . In accordance with embodiments of the present invention, the current signal includes one or more modulated signals on a carrier signal. In some embodiments, the current signal generated by transmitter  101  can include signals at more than one carrier frequency in order that the influences associated with coupled neighboring conductors, which can be frequency dependent, can be more clearly distinguished. 
   In a typical locate scenario, other lines adjacent to target line  104  may be present underground. Depending on the physical layout, these lines can also form part of an AC circuit that carries direct or coupled currents from transmitter  101 . As shown in  FIG. 1B , through a shared ground connection at splice box  102 , line  105  can be electrically coupled to target conductor  104 . As illustrated here, conductor  105  is resistively coupled to target conductor  104 , i.e. the current moving through target conductor  104  uses line  103  as a lower impedance ground return path to transmitter  101 . 
   Another current that is also a result of the signal generated by transmitter  101  is flowing in conductor  106 . This current is coupled into conductor  106  by virtue of inductive or capacitive coupling from target line  104  to conductor  106 . In effect, a separate current loop is set up through a loosely coupled ground  108 , so that the sense of the signal in coupled conductor  105  is still positive, i.e., in the same direction as the target conductor. 
   A line locator measuring the signal strength in region  103  thus sees a combination of signals due to the magnetic fields emanating from conductors  104 ,  105 , and  106 . The sum of these signals is a distorted field, since it no longer is concentric around the axis of targeted line  104 , as it would be if only targeted line  104  were carrying a current signal at the locate frequency. Some locators have been able to discriminate the reverse direction signal  105  from the forward signal on the targeted line  104 . Embodiments of the present invention can also explicitly identify the coupled signal present on conductor  106 . 
     FIG. 2  shows a simplified equivalent circuit to that shown in  FIG. 1B . In particular,  FIG. 2  illustrates how mutual induction generates an AC current in coupled line  106 . As shown in  FIG. 2 , the ground paths are loosely coupled as well.  FIG. 2  is a simplified schematic of the situation, showing only the coupled conductor  106  carrying current I i  and the target conductor  104  carrying current I g  (for the galvanic, or directly coupled current). Transmitter  101  is represented in this schematic as a signal generator. 
     FIG. 3A  is a further reduction of the problem to an equivalent electrical schematic, with the loops  104  and  105  coupled together through mutual inductance  301 . It is well know that in the case of inductive coupling, the induced current I i  lags the current in the primary loop I g  by 90°, as is shown graphically in  FIG. 3B . 
   Note that if the coupling between conductors  104  and  106  were capacitive instead of inductive, the situation is very similar. In this case, however, the current in the neighboring conductor  106  will lead the current in the primary loop target conductor  104  by 90°. 
   In accordance with embodiments of the present invention, the current signal applied to target conductor  104  ( FIG. 1 ) by transmitter  101  includes a carrier frequency, which is the active line locate frequency discussed above. Further, a FM modulation frequency can be imposed on the current signal. In some embodiments, the FM modulation frequency is an integer multiple of the carrier frequency. 
     FIG. 4  illustrates a block diagram of a receiver  410  according to the present invention. As discussed above, receiver  410  is included in locator  110 . Receiver  410  includes a detector  409  that detects the strength of an electromagnetic field and provides one or more input signals  401 . In some embodiments, detector  409  can include magnetic field detectors as well as filters and digitizers. In some embodiments, locator  410  can include multiple individual receivers  410  with detectors  409  oriented to detect magnetic fields of the electromagnetic field that are directed in particular directions. Receiver  410  locks first onto the carrier frequency, i.e. the active locate frequency, in a digital phase-locked loop DPLL  402  and then onto the FM modulation frequency in DPLL  404  to provide a signal (the target line strength) that is not substantially influenced by inductive or capacitive bleedover to neighboring conductors. 
   In receiver  410 , then, processing first demodulates the Signal Select FM signal from the carrier signal, and then subsequently demodulates the original input signal by the detected reference phase. Input signal  401 , then, is first received in carrier DPLL  402 . The output signal from DPLL  402  is coupled into FM Demodulation DPLL  404 . Therefore, DPLLs  402  and  404  lock first to the carrier frequency, and next to the FM modulation frequency, which is an error term resulting from operation of the DPLL  402 . In some embodiments, a downsampler  403  can be provided between DPLL  402  and DPLL  404  to improve processing efficiency because the FM modulation frequency can be a fraction of the carrier frequency. Some embodiments of carrier DPLL  402 , downsampler  403 , and FM demodulation DPLL  404  are further discussed in the &#39;376 application. 
   An output signal from DPLL  404  is coupled into quadrature demodulator  406 , where it is utilized to demodulate input signal  401 . The resulting output signals from quadrature demodulator  406  include an in-phase signal  408  that is the target line signal strength and a quadrature signal  407  that is the inductive or capacitive bleedover line signal strength. In some embodiments, a quadrature upsampler  405  can be included so that the sampling rate of in-phase signal  408  and quadrature signal  407  is adjusted. Signals generated in receiver  410  can be further processed for display to user  116  and display  114 . 
     FIG. 5  illustrates an embodiment of carrier DPLL  402  according to the present invention. Carrier DPLL  402  includes a mixer  501  that mixes signal input  401  with a periodic function generated by numerically controlled oscillator (NCO)  503 . A phase error signal  504  is created by inverse tangent function  502  that receives a complex output signal from mixer  501 . A simple divide of the imaginary term by the real term of the input signal can also form a simple estimate of the phase error, since the inverse tangent function  502  is approximately linear about zero. The phase error signal  504  from inverse tangent function  502  is input to NCO  503  that adjusts the phase and frequency of the periodic function output to mixer  501  and can also output values for the phase and frequency. In some embodiments, NCO  503  can use second order loop equations to control the feedback to mixer  501 , allowing DPLL  402  to gradually converge to a lock condition. At the lock condition, signal  505  represents the difference between the signal input  401  and the average carrier at frequency F c , and is taken as a representation of the FM signal and is forwarded to FM DPLL  404 . 
     FIG. 6  illustrates another embodiment of receiver  410  according to the present invention. As shown, input signal  401  is input to quadrature demodulator  406  and carrier DPLL  402 . As was discussed with regard to  FIG. 5 , carrier DPLL  402  includes mixer  501 , arctangent function  502 , and NCO  503 . The FM signal from carrier DPLL  402  is input to FM DPLL  404 . Again, as in DPLL  402  shown in  FIG. 5 , FM DPLL  404  includes a mixer  611 , an arctangent function block  612 , and a numeric controlled oscillator  613 . As shown in  FIG. 6 , mixer  611  receives the FM signal from DPLL  402  and mixes an FM modulated signal output from numerically controlled oscillator  613 . Numerically controlled oscillator (NCO)  613  adjusts the frequency and phase of the periodic function mixed in mixer  611  according to the FM phase error signal generated by arctangent function  612 . In some embodiments, NCO  613  can use second order loop equations to control the feedback to mixer  611 , allowing DPLL  404  to converge to a locked condition. The periodic feedback function input to mixer  611  in DPLL  404  is also input to quadrature demodulator  406 . 
   In some embodiments, the magnitude  601  of the complex FM signal output from DPLL  402  can be input to filter  602  to provide a total signal strength signal  603 . Total signal strength signal  603  includes contributions from target conductor  104 , resistively coupled neighboring conductors such as conductor  105  shown in  FIG. 1B , and inductively coupled neighboring conductors such as conductor  106  shown in  FIG. 1B . 
   As shown in  FIG. 6 , quadrature demodulator includes a mixer  614 , an in-phase filter  616 , and a quadrature filter  615 . Mixer  614  mixes input signal  401  with the sinusoidal function generated by DPLL  404 . The output signal from mixer  614 , then, is a DC signal with a real and imaginary portion. The real, or in-phase, portion is a result of signals originating from target conductor  104  or from conductors that are resistively coupled to target conductor  104 . Therefore, in-phase filter  616  isolates the real portion of the signal output from mixer  614 . The output signal from in-phase filter  616  can be input to filter  604 . The output signal from filter  604  indicates the signal strength from target conductor  104  and from conductors that may be resistively coupled to target conductor  104 . The sign of the output signal from filter  604 , as is discussed in the &#39;136 application, indicates the signal direction. The sign can be determined in sign block  605  in order to provide a signal direction signal. 
   The quadrature portion of the output signal from mixer  614  can be isolated in quad filter  615  and filtered in filter  617 . The output signal from filter  617  provides the signal strength due to signals detected from neighboring conductors that are inductively or capacitavely coupled to target conductor  104 . 
   Therefore, mixing the detected FM signal in quadrature with the incoming signal  401  allows for isolation of contributions from targeted conductor  104  and those conductors that may be resistively coupled to targeted conductor  104  from signals that are due to conductors that are capacitively or inductively coupled to targeted conductor  104 . As is shown in  FIG. 4 , in some embodiments quadrature upsampler  405  can be included in order to return the sample rate back to that of input signal  401 . 
   As is illustrated in  FIG. 6 , the phase reference established by transmitter  101  between the carrier signal and the FM signal is recovered by the combination of DPLLs  402  and  404 , and this can be used to decouple the bleedover signal strength from the signal strength due to targeted line  104  and other resistively coupled elements. A quadrature demodulator is well suited for this task, representing a multiplication by a complex sinusoid, and results in the decomposition of the input signal into its real and imaginary parts. The imaginary part corresponds to the inductively or capacitively coupled signals that are 90° out of phase with respect to the real part. The real part represents the resistively coupled signals emanating from the target conductor and those conductors carrying return currents in the reverse direction. 
   The embodiment of receiver  610  shown in  FIG. 6  represents an embodiment of a signal processing system that simultaneous calculates total signal strength  603 , target line signal strength  606  (the real part of the quadrature demodulator  406 ), and the signal direction. Some embodiments may not include determination of non-resistive bleedover signal strength determination because the primary aim of receiver  610  may be to provide a signal strength estimate  606  that is cleaned of distortion, and thus can serve as the basis for unbiased depth and current measurements. The sign of the target line strength signal output from filter  604  is the signal direction of the resistively coupled component of input signal  401 , which is used to determine whether the target signal is outgoing from the transmitter, or a ground return current such as that present on line  105 . In this topology, the signal direction parameter is filtered and thus represents a true average; unlike more conventional methods that require phase comparisons between a reference carrier signal and a modulated signal. Filters  602 ,  604 , and  605  can have low-pass characteristics and may be designed to dampen the signal strength values presented to the user via display device  114  ( FIG. 1A ). 
   The total signal strength measurement  603  is what is traditionally provided by line location systems. As has often been noted in the art, signal distortion present in signal  603  can lead to biased estimates of target line depth and position. In some embodiments, the relative amount of field distortion can be determined by a comparison of the target line signal strength signal output from filter  604  and the total signal strength signal output from filter  602 . When total signal strength signal  603  is substantially the same as the target line signal strength output from filter  604 , the measurement can be said to be free of distortion. 
   With the target line signal strength signal from filter  604  representing the field strength of target conductor  104  being cleaned of the effect of signal bleedover due to inductive or capacitive coupling, it is possible to use an optimization algorithm to compute the unbiased depth of target conductor  104  and current in target conductor  104  without substantial risk of converging on local minima, which might be the case if the distorted total signal strength signal  603  were used in the calculation. The above discussed signals generated in receiver  610  can be input to display  620 . Display  620  can include a processor  622  and a user interface  624 . Processor  622  can calculate various parameters and results based on, for example, the signal direction and target line signal strength signals. User interface  624  then can display results to user  116 . 
   For example,  FIG. 7  illustrates the geometry for determining the depth of target conductor  104 . With the current I g  on target conductor  104  flowing out of the paper, the user establishes an arbitrary reference position  701  and walks perpendicularly across the line in direction  702 . There are three unknowns: the current I g , the centerline position x 1 , and the depth z. Several to many independent measurements of the target line signal strength signal from filter  604  are collected automatically as the user walks across the line by, for example, processor  622  in display  620 . In  FIG. 7 , the x-position of these measurements are noted by solid marks along the ground surface, and it is obvious to those trained in the art of signal processing that the x-increment between these measurements can be arbitrarily small and is limited only by the available instruction cycles. 
   Assuming a long linear target conductor  104 , the signal strength amplitude for a distortion-free signal is given by the equation: 
                   h   n     :=       I     [     2   ·   π   ·           (       x   n     -   x1     )     2     +     z   2           ]       ·     (     cos   ⁡     (     atan   ⁡     (         x   n     -   x1     z     )       )       )               (     EQN   .           ⁢   1     )               
where I g  is the amplitude of the unknown current in the cable, which is constant for all measurements h n . Note that for cylindrical fields, h n  is inversely proportional to the radius from the measurement point to the cable (as is shown in the equation for x n =x 1 ). Only the series of amplitude measurements along the ground in the x-direction (at known x-intervals) is known. For simplicity, we can assume that coordinate x is perpendicular to the cable axis, although in some embodiments the analysis can be expanded such that field strength measurements may be taken at a set of arbitrary x, y, and z positions (where y is the direction of the cable). In some embodiments, the x, y, and z positions can be determined by a position determination system included in locator  110 .
 
     FIG. 8A  illustrates two buried cables  104  and  106  separated by an unknown distance. The solid line represents the galvanic current in the resistively coupled target conductor. The dashed line represents the induced current in a parallel line, which may be at a different depth underground.  FIG. 8A  also shows the key measurement sensors in an embodiment handheld locator  110  that allow an automatic creation of the x-vector corrected field strength plots shown in  FIG. 8B . 
     FIG. 8B  shows the field strength magnitude (dotted line), as well as the inphase (dashed line) and quadrature (solid line) component outputs that result from the bleedover decoupling method. The field strength magnitude is composed of signals from both the target conductor as well as a parallel conductor that is carrying a current due to mutual inductance. Estimation errors will result if this signal strength magnitude is used to vector the depth and position of the target line. 
   As shown in  FIG. 8A , handheld line locator  110  with receiver  410  according to the present invention are coupled to display  620 , which can include processor  622  and user interface  624 , to display the total field strength from which the user can deduce the presence of a target conductor  104 , in rough coordinates. However, the total field strength signal is distorted by the presence of conductor  106 , which carries current I i  in the same direction as conductor  104  due to inductive bleedover. Lines  104  and  106  are separated by an unknown distance Δx. The bleedover decoupling method as implemented in the signal processing receiver  410  shown in  FIG. 6  allows a separation of the signals emanating from conductors  104  and  106  (and measured by reference coil  804 ), thus allowing an improvement in the estimate of location of target conductor  104 . 
   In addition to the reference coil  804  from which is derived the input signal  401  in detector  409 , locator receiver  110  can include a 3-axis inertial position tracking system  802 , an electronic Earth compass  803 , and a guidance coil  805 . With these devices, each of which can provide measurements at small intervals x, it is possible to convert a walking traverse of the line in a generally transverse direction to an exactly perpendicular traverse along a line calibrated in distance units. Earth compass  803  provides a general heading that should be followed by the user, and is used to correct the walked line back to a straight path. The inertial position (as derived by integrations of the 3-axis acceleration in trading  802 ) allows the ability of the system processor to calculate absolute distance along the path (in 3-d space). In the concentric field of a long linear conductor, the guidance coil will have a null response when the walking line is exactly perpendicular. Thus the measured field strength of the guidance coil can be used to correct the walking path to a perpendicular across the target pipe or cable. 
   Clearly other such means of correcting the measurement position to a perpendicular path across the target cable are possible, using laser rangefinders, radar or ultrasonic position measurement devices, and simple measuring tools like wheels and tape measures. 
   In this way, the position of reference coil  804  during the “walkover locate” can be corrected to allow a presentation of the field strength as shown in  FIG. 8B . The position along the abscissa x represents the perpendicular distance across the line, with an arbitrary starting position X ref . Trace  808  is the overall magnitude of signal strength as derived from a recorded set of signals  603 . Trace  809  represents the substantially distortion free signal from the inphase output of the quadrature demodulator, as derived from a recorded set of signals  606 . It is negative, since by convention the positive direction y refers to the target line direction, and in the example presented in  FIG. 8A , the current is flowing in the negative y direction. At each measurement point on the x-axis, the sign of the filtered inphase output  605  of quadrature demodulator  406  is taken as the signal direction. For comparison, trace  807  is the filtered quadrature output as a function of transverse position over the line. Trace  807 , then, represents the signal emanating from the coupled conductor  106 , which is 90° out of phase with respect to that from target line  104 . 
   The simultaneous solution for z, x 1 , and I g  is a nonlinear minimum mean square problem. One embodiment of the solution is to use an optimization algorithm, for example a Levenberg-Marquardt algorithm, and to fit the data in three dimensions simultaneously (unknown x 1 , z, and I). Initial conditions of all three quantities are deduced based on rough field information. The Levenberg-Marquardt algorithm uses multiple iterations to minimize the sum of squares of the quantities |h n −h n ′|, where h n ′ are the measured values of the signal strength along the x axis. 
   If the set of signal strength results  808  (representing the total field strength from both conductors) are taken as the measurement values h n ′ in EQN. 1, there will be a biased outcome, as indicated by the centerline error  810 . An even worse behavior is that the optimization will converge to the wrong set of values due to the distortion in the 1/r field as a result of the presence of the coupled conductor. But when the target line field strength result set  809  forms the input to the optimization routine, a well-behaved simultaneous solution for x 1 , z, and I g  can be obtained and can have good convergence properties. In this case no bias is introduced in the centerline or depth position. 
   The embodiments described herein are examples only of the invention. Other embodiments of the invention that are within the scope and spirit of this disclosure will be apparent to those skilled in the art from consideration of the specification and practice of the invention disclosed herein. It is intended that the specification and examples be considered as exemplary only and not limiting. The scope of the invention, therefore, is limited only by the following claims.