Abstract:
An example wireless device includes a radio receiver to measure a signal quality of a data signal independent of a direct frequency measurement, the signal quality correlated to an offset between a transmitter reference frequency and a receiver reference frequency but not indicative of a direction of the offset. The example wireless device further includes a reference frequency generator to determine from the measured signal quality that a previous adjustment to the receiver reference frequency in a first direction has worsened the signal quality, and responsive to that determination adjust the receiver reference frequency in a second direction that is opposite to the first direction.

Description:
RELATED APPLICATIONS 
     This application is a continuation of U.S. patent application Ser. No. 12,871,814, filed Aug. 30, 2010, which is a continuation of U.S. patent application Ser. No. 11/006,998, filed Dec. 7, 2004, now U.S. Pat. No. 7,787,829, issued Aug. 31, 2010, which claims priority to U.S. Provisional Application No. 60/532,753, filed Dec. 23, 2003, each of which are incorporated herein by reference in their entirety. 
    
    
     TECHNICAL FIELD 
     The present invention relates generally to electronic circuits and in particular to circuits used for data communications. 
     BACKGROUND OF THE INVENTION 
     Conventional short range data radios typically divide the radio spectrum within which they operate into non-overlapping frequency channels. For example, radios with a 1 Megahertz (MHz) occupied bandwidth operating in the 2.4 Gigahertz (GHz) Industrial Scientific Medical (ISM) band typically divide that spectrum into approximately eighty 1-MHz wide channels. These radio systems transmit and receive data using frequency modulated Radio Frequency (RF) signals centered on one of these 1-MHz channels. In some cases, the transmitter may hop between channels during normal data transmission. In other cases, having found a good channel, the transmitter may continue to use that one channel unless or until data transfer on that channel becomes unreliable. 
     Typically, these radio systems generate the RF carrier frequency by multiplying the frequency of a low frequency crystal oscillator up to the RF frequency used for transmission. Many 2.4 GHz radio systems use 13 MHz crystals for this purpose, but crystal frequencies in the 12-32 MHz range are also common. 
     Radio receivers, especially Frequency Modulation (FM) receivers using a low Intermediate Frequency (IF), typically implement a Band Pass Filter (BPF), through which the mixed-down signal is passed before demodulation. This is necessary in order to prevent RF signals on adjacent channels from being demodulated, or interfering with the reception of signals on the channel the receiver is currently configured to receive. 
     The crystal oscillators of both the transmitter and receiver should be oscillating at almost exactly the same frequency. If not, part of the transmitted signal may be attenuated by the receiver&#39;s BPF. This is shown in  FIG. 1 . Frequency response  12  shows the frequency spectrum of a transmit signal  18  and the frequency operation of a receiver&#39;s Band Pass Filter (BPF)  16 . When the transmitter and receiver have crystal oscillators with the same frequency, the transmit signal  18  should be substantially centered within the BPF  16 . In this common reference frequency condition, the transmitted signal  18  will have minimum attenuation. 
     Frequency response  14  shows the frequency spectrum when the transmitter and receiver have crystal oscillators with different (offset) reference frequencies. In this offset frequency situation, the transmit signal  18  is no longer centered within the BPF  16 . Any portion of the transmit signal  18  extending outside of BPF  16  is attenuated, such as the shaded portion  20 . The attenuation  20  lowers the signal strength of signal  18  and can prevent the receiver from successfully or reliability receiving data carried in the transmit signal  18 . 
     In one example, a radio system may operate at 2.4 GHz, with a 1 MHz channel spacing and a 900 kHz occupied bandwidth. At 2450 MHz, a 50 parts per million (ppm) offset is equal to 122.5 kHz. As the occupied bandwidth is 100 kHz less than the channel spacing, there is 50 kHz on either side of a perfectly centered transmitter spectrum that is not part of the adjacent channel. In the 50 ppm offset example, 7.2% of the transmitted signal extends into the adjacent channel. Typically, the receiver BPF is a little wider than the channel, and the roll-off of the filter is not a “brick wall”, so a small offset can be tolerated with minimal impact on receive sensitivity. In a typical 2.4 GHz radio system, a 50 ppm offset is approximately the maximum that can be tolerated without significantly impacting performance. 
     Conventional wireless solutions use quartz crystals to derive a radio carrier reference frequency. These conventional solutions have disadvantages, including requiring expensive, high accuracy crystals. Even using such crystals, significant offsets may exist between the transmit and receive frequency resulting in reduced receive sensitivity. Even moderately affordable crystals may require time-of-manufacture crystal trimming, thereby increasing manufacturing cost and complexity. 
     The receiver and transmitter are each subject to separate oscillator frequency inaccuracies. Therefore the receiver and transmitter require a crystal with an accuracy of better than +/−25 parts per million (ppm) to prevent the oscillator accuracy from impacting system performance with a combined worst case error of greater than 50 ppm. 
     Crystal oscillator accuracy is typically specified as three components; initial tolerance, temperature variation, and long-term drift. In order to put products in the best possible light, crystal oscillator vendors typically quote only the initial tolerance. Frequency variation with temperature is usually similar to the initial tolerance, and aging is usually in the range of one to five ppm per year. 
     Crystals generally drift in the same direction. Two instances of the same crystal would not usually drift in opposite directions, but they may well drift at different rates in the same direction. One factor affecting drift is the drive strength of the oscillator circuit driving the crystal. Another factor may be the proportion of time that the crystal oscillator is active. Over 5 years, a crystal with a 3 ppm/year drift spec may drift only 5 ppm (or less) or not at all, while another may drift 15 ppm. This crystal drift depends on drive strength, the amount of time that the oscillator is running, and the physical properties of the individual crystal. 
     The frequency variation with temperature is not linear, but rather typically a quadratic or cubic curve. Therefore, variations in temperature across only part of the rated range may cause frequency to vary over most of the stated tolerance. 
     Therefore, a 25 ppm crystal, which initially may appear to be suitable for uses in the wireless applications discussed above, may not in fact be suitable. Such a crystal would typically have 25 ppm initial tolerance, 25 ppm variation over temperature, and 3 ppm/year drift. One such crystal, starting at 25 ppm, and operated at a temperature which caused the frequency to oscillate at close to its minimum frequency, may be oscillating at minus 50 ppm from its nominal after 5 years. Another crystal, starting at +25 ppm, at a different temperature may oscillate at +65 ppm from its nominal after 5 years, resulting in a difference of 115 ppm. This drift could severely impact the receive sensitivity of almost any 2.4 GHz radio system. Therefore, to meet the +/−25 ppm spec discussed above, a more expensive 10 ppm crystal is required. 
     Above 30 ppm, the cost savings from specifying a looser tolerance is low. For example, one vendor may offer a 13 MHz 30 ppm crystal in volume at a given price, and the 50 ppm version of the same crystal may only be a few cents cheaper. However, crystals with tolerances below 30 ppm quickly become more expensive, and a 10 ppm crystal may typically cost 3× to 5× more than the 30 ppm crystal. Frequency accuracy requirements of a design may therefore place a significant cost burden on low cost wireless systems. 
     For this reason, many low cost radio Integrated Circuits (ICs) include a feature allowing trimming of the initial crystal frequency. Typically, this is implemented by using a digitally trimmable capacitance. At manufacturing test of a wireless product, the oscillator frequency is measured, and an appropriate trim factor is stored in non-volatile memory within the device. This trim factor is loaded into a radio Integrated Circuit (IC) after each reset and allows the digitally controlled capacitance of the crystal oscillator to tune the crystal frequency to a nominal value. This removes the initial tolerance component of the oscillator, reducing the variation to just the temperature and drift components. 
     This technique allows the use of 15 ppm crystals with many 2.4 GHz radio ICs, without impacting radio performance. However, this comes at the cost and trouble of implementing crystal tuning during manufacture. Regardless, 15 ppm, and even 20 ppm, crystals are still much more costly than 30 ppm crystals. 
     It would be desirable to use much less accurate crystals in low cost radio transmitters and receivers. 
     SUMMARY OF THE INVENTION 
     Wireless devices transmit and receive radio signals based upon reference frequencies that are generated by reference frequency circuits. If the reference frequency in the transmitter is different from the reference frequency in the receiver, the radio signals may not be received properly or may not be capable of being received at all. A measurement circuit measures the amount of error or signal corruption in radio signals due to the reference frequency offset. A trimming circuit then tunes the reference frequency in the transmitter or receiver to reduce the reference frequency offset. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows attenuation of a transmitted signal when the transmitter and receiver reference frequencies are offset. 
         FIG. 2  shows an exemplary implementation of a system that provides reference frequency compensation. 
         FIGS. 3 and 4  show one example of how the reference frequency is compensated in  FIG. 2 . 
         FIG. 5  shows a radio device that provides reference frequency compensation. 
         FIG. 6  is a detailed diagram of a crystal oscillator frequency trim circuit that can be used in the radio shown in  FIG. 5 . 
         FIG. 7  is one example of radio transmit circuitry that provides reference frequency compensation. 
         FIG. 8  is one example of radio receive circuitry that provides reference frequency compensation. 
         FIG. 9  is a block diagram showing how reference frequency compensation can alternatively be performed within a frequency synthesizer. 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 2  shows reference frequency tuning system  21  that tunes the reference frequency of a radio receiver  34  with the reference frequency of a radio transmitter  22 . The radio transmitter  22  includes an antenna  26  that transmits signals  28  that are received by the antenna  32  of receiver  34 . In one embodiment, the radio receiver  34  calculates signal quality information  33  associated with the received signal  28  and then uses that signal quality information  33  to adjust the reference frequency  38  used by the radio receiver  34  for demodulating the transmit signal  28 . For example, a trimming operation is performed in the reference frequency generation circuit  36  that tunes the output frequency  38  to the frequency  25  used by the reference frequency circuit  24  in the transmitter  22 . In other words, the reference frequency  38  in the receiver  34  is varied until it is tuned (zero offset) with the reference frequency  25  in the transmitter  22 . 
     Signal quality information  33  can be any signal measurement or statistic that indicates the transmitter  22  and the receiver  34  have offset reference frequencies. For example, the signal quality information  33  may be a Bit Error Rate (BER), packet error rate, signal strength, or any other indicator of frequency offset. Calculating these signal quality measurements have been used for other purposes in the past, and are therefore known to those skilled in the art, and are therefore not described in further detail. 
     In an alternative embodiment applicable in two-way systems, the receiver  34  may send the signal quality measurements  33  back to the transmitter  22  in signal  30 . The transmitter  22  then trims its reference frequency  25  according to the signal quality measurements  33  in signal  30 . In another embodiment, the signal  30  may just carry the ppm value for the desired frequency offset to use for reference frequency  25 . The transmitter  22  varies the reference frequency  25  until the signal quality information  33  identified in reply signal  30  is within an acceptable level. 
     The tuning system  21  may use a single frequency in normal operation (one of many channels available) or may use a frequency hopping system such as used in Frequency Hopping Spread Spectrum (FHSS) systems. The tuning system  21  can also use a single point to point (1-1) radio link or can be implemented in a network of wireless devices, with an overall network coordinator or master. 
     It is not essential that the transmitter  22  and receiver  34  operate exactly on a pre-determined nominal frequency. For example, the transmitter  22  and receiver  34  may be designed to operate at a nominal frequency of 2450 MHz. The radios can work equally as well if both are centered at 2450.1 MHz. There may be a slight reduction in immunity to adjacent channel signs, but in general, the sloped (as opposed to brickwall) roll-off of the receiver BPF means that wireless systems typically try to avoid using adjacent channels between 2 pairs of closely located devices. 
     If the transmitter and receiver frequencies are significantly offset, even by one or two hundred ppm, the receiver  34  will not completely fail to receive the transmitted signal. Instead, the Bit Error rate (BER) on the wireless link  28  will be very high. This will often make it difficult for the receiver  34  to successfully receive significant quantities of transmitted data, but the receiver  34  will still be aware that the signal  28  is being transmitted. Sufficient data will get though so that the receiver  34  will be able to recognize the transmitter  22  as its counterpart, rather than an interfering signal. 
     In the case of two-way wireless systems, where each wireless device includes both a radio transmitter and receiver, only one of the wireless devices might incorporate the reference frequency trimming operation. In a many to 1 wireless radio system, such as a system in which multiple “slaves” send data to a single “master” receiver, the frequency trimming operation might be incorporated in each of the slaves, which would then seek to tune their radios to the master device. 
       FIGS. 3 and 4  show one example of how the reference frequency is turned in the reference frequency circuits  24  or  36  in  FIG. 2 . The tuning operations described in  FIGS. 3 and 4  can be performed in the receiver  34  ( FIG. 2 ) or can be performed in the transmitter  22  ( FIG. 2 ). For illustrative purposes, the description in  FIGS. 3 and 4  will refer to the frequency trimming operation as being performed in the receiver  34 . 
     The transmitter  22  ( FIG. 2 ) sends data to the receiver  34  and the receiver  34  demodulate the received data in block  40  of  FIG. 3 . The data may be successfully received. For example, there may be sufficiently few bit errors after error correction techniques have corrected the data. The receiver in block  40  accordingly sends back an acknowledge handshake packet to the transmitter and the transaction completes. 
     However, the bit error rate in the data may be uncorrectable. In this case, the receiver  34  will not respond. After a timeout period, the transmitter  22  will then retransmit the data packet. Having received a corrupted data packet, the receiver in block  44  stores a measure of the bit error rate of the corrupted packet. The receiver then changes its oscillator trim capacitance attempting to correct for an inferred oscillator frequency offset between the transmitter and receiver. An example of the frequency offset between the transmitter  22  and receiver  34  is represented by initial frequency response  60 A in  FIG. 4 . The initial frequency response  60 A shows the transmit signal  28  offset from the expected frequency response of the Band Pass Filter (BPF)  59  of the receiver  34 . This is similar to the frequency response  14  shown in  FIG. 1 . 
     The initial offset frequency response  60 A results in the transmit signal  28  generating a high BER in block  44  of  FIG. 3 . If a first trim operation is being performed in block  46 , the receiver shifts its oscillator frequency by 25 ppm in a random direction in block  48 . This of course is just one example and other shift values can also be used. In this example, the receiver reference frequency is shifted +25 ppm and is shown graphically as first trim stage  60 B in  FIG. 4 . The receiver  34  then waits for the transmitter  22  to transmit another packet. 
     The initial +25 ppm trim in the first trim stage may have provided the desired result by reducing the BER below some threshold value in block  44  of  FIG. 3 . Accordingly, the receiver  34  sends an acknowledge to the transmitter  22  in block  42  confirming to the transmitter  22  that the receiver  34  is now operating at the same, or substantially the same, oscillator frequency. The receiver  34  stores the trim value and continues to use that oscillator trim setting until a future uncorrectable packet is received. 
     However, the BER in block  44  may still be above some BER threshold after the first trim stage  60 B. During a second trim stage in block  50 , the receiver  34  first compares the BER of first trim stage  60 B with the initial signal response  60 A ( FIG. 4 ). If the first trim stage  60 B improved the BER in block  52 , the receiver  34  infers that the oscillator frequency was trimmed in the correct direction, but not far enough. In this case, the receiver  34  trims its crystal another 25 ppm in the same direction in block  48 . 
     However, the first trim stage  60 B may make the BER worse than initial signal response  60 A. This is the case in  FIG. 4  where the first trim stage  60 B moves the transmit signal  28  further outside of BPF  59 . In this case, the receiver  34  infers that the oscillator frequency was trimmed in the wrong direction and trims the oscillator frequency 50 ppm in the opposite direction in block  54 . This is 25 ppm on the other side of the original frequency and is shown as second trim stage  60 C in  FIG. 4 . 
     The process is repeated. In the example shown in  FIG. 4 , the BER rate is still above the BER threshold value in block  44  after the second trim stage  60 C. However, in block  56 , the BER after the second trim stage  60 C is better then the BER of the previous trim stage  60 B. Accordingly, in block  57  the receiver  34  trims the oscillator frequency another 25 ppm in the same direction as the previous trim stage. This is shown as third trim stage  60 D in  FIG. 4  that shifts the reference frequency in the receiver another 25 ppm. 
     While showing improvement, the BER rate for the third trim stage  60 D is still above the BER rate threshold in block  44 . The receiver in block  57  again trims the crystal oscillator frequency another 25 ppm in block  57 . This is shown as the fourth trim stage  60 E in  FIG. 4 . After the fourth trim stage  60 E, the BER rate is now within an acceptable range in block  44 . The receiver  34  accordingly sends the acknowledge signal to the transmitter  22  in block  42 . 
     As shown above, the process above iteratively repeats until a packet is either correctly received, or the transmitter times out. A timeout may occur due to the presence of an on-band interfering signal, or because the transmitter and receiver are further apart in distance than the operating range of the wireless link can support. 
     In many cases, a significant bit error rate may exist in normal communications, especially towards the limits of distance range. Uncorrectable packets may therefore be received for a reason other than reference frequency offset. In this case, the receiver would not try shifting its oscillator trim value until a number of successive uncorrectable packets had been received. 
     Transmit and Receive Circuitry 
       FIG. 5  shows in more detail an example of circuitry that may exist in the transmitter  22  or receiver  34  for trimming a reference frequency. An antenna  66  receives or transmits wireless signals. A radio transceiver  68  is coupled to the antenna  66  and communicates to a microcontroller  62  through a logic interface  70 . The untrimmed reference frequency  75  for the radio transceiver  68  is generated by a crystal oscillator  74 . A crystal oscillator frequency trim circuit  72  varies or “trims” the untrimmed base frequency of the crystal oscillator  74  and outputs the trimmed frequency  75  to the radio transceiver  68 . Adjustment of the base frequency of the crystal oscillator  74  can be controlled by logic in the radio transceiver  68  or by the microcontroller  62 . 
     Over time, temperature, etc. the original specified frequency crystal oscillator  74  may change. As described above, the frequency  75  is varied by trim circuit  72  as described above in  FIGS. 3 and 4  to compensate for the frequency offset between the transmitter and receiver. 
       FIG. 6  shows one example of the crystal oscillator frequency trim circuit  72  in  FIG. 5 . Of course, any circuit that varies a crystal oscillator frequency can be used, and the circuit in  FIG. 6  is only one example. An amplifier  82  with hysteresis is coupled across opposite ends of the crystal oscillator  74 . A set of capacitors  84  are selectively coupled to the amplifier  82  by switches that are activated by signals  80 . In one example, the signals  80  are generated by the microcontroller  62  ( FIG. 5 ) through the logic interface  70 . The capacitors  84  are selectively connected or disconnected from the amplifier  82  to vary the base frequency of the crystal oscillator  74 . The varied frequency  75  is output to the radio transceiver  68 . 
       FIG. 7  shows one example of the functions that may be contained in the radio transceiver  68  in  FIG. 5  or inside the transmitter  22  shown in  FIG. 2 . Again, this is only an example and any type of transmitter circuitry can be used. In this example, a frequency synthesizer  86  receives the trimmed crystal oscillator frequency  75  from the trim circuit  72 . 
     The frequency synthesizer  86  includes a phase detector ( FIG. 9 ) that receives the output  75  of the crystal oscillator trim circuit  72  and outputs a signal to a Voltage Controlled Oscillator (VCO) ( FIG. 9 ). The output frequency  87  is equal to the crystal oscillator frequency  75  multiplied by a devisor in a feedback loop. Typically, the divider is used to set the operating channel for the transmitter. 
     The frequency synthesizer  86  generates output carrier frequency  88  that is mixed with the transmit data  92  by a mixer  91 . The output from the mixer  91  is amplified by an amplifier  90  and output over antenna  66 . The microprocessor  62  is connected to the amplifier  90  through logic interface  70 . 
       FIG. 8  shows a diagram of the receive circuitry that may exist inside the radio transceiver  68  or inside the receiver  34  shown in  FIG. 2 . The trim circuit  72  again feeds the trimmed crystal oscillator frequency  75  to a frequency synthesizer  94  similar to frequency synthesizer  86  in  FIG. 7 . The output frequency generated from the synthesizer  94  is mixed with the signal received over antenna  66  by mixer  96 . The output of the mixer  96  is filtered by a Band Pass Filter (BPF)  59  as shown in the example of  FIG. 4 . The output of the BPF  59  is FM demodulated by demodulator  102  and output as receive data  104 . 
     Alternative Embodiments 
       FIG. 9  shows one alternative embodiment where the crystal oscillator frequency is trimmed by varying the output frequency of the frequency synthesizer  86  or  94  in  FIG. 7  or  8 , respectively. A programmable frequency divider  114  is coupled between a phase detector  110  and the output of a VCO  112  that varies the output frequency  87  of the frequency synthesizer  86  or  94 . Instead of trimming the output of the crystal oscillator  74 , the microcontroller  62  varies the output frequency of the synthesizer  86  or  94 . The output frequency  87  is varied to again compensate for frequency offset between the transmitter and receiver. 
     The frequency of output  87  is varied by the microcontroller  62  by digitally varying the value in the frequency divider  114 . The digital value of the frequency divider  114  can vary the frequency of the output signal  87  or  88  in both a positive ppm and negative ppm direction. The same iterative process shown in  FIGS. 3 and 4  is performed where the value of the frequency of output signal  87  or  88  is repeatedly varied by the microcontroller  62  until the BER or signal strength of the received data signal is within some acceptable value. 
     Referring back to  FIG. 8 , in some cases, it may not be possible to readily determine the BER of the received packet, but merely that it contains errors. This may occur for example because a checksum or Cyclic Redundancy Check (CRC) byte does not match the data. In this case, an alternative way of judging relative effectiveness of varying frequency trim values is to use a Received Signal Strength Indicator (RSSI) circuit  98 . The RSSI circuit  98  may already be found in many radio receiver ICs. 
     The RSSI  98  is typically used to detect relative differences of as little as 1 dB in signal strength reliably. The RSSI  98  is typically measured after the receiver band pass filter (BPF)  59 . Thus, for a given distance between transmitter and receiver, and a given physical environment, the relative signal strength indicated by the RSSI  98  is a good indication of the relative closeness of the receiver oscillator trim frequency  75  to the oscillator frequency in the transmitter. The crystal oscillator trim values are varied until the RSSI is at the maximum value achievable within the oscillator trimming range. 
     Training Sessions 
     In some systems, it may not be acceptable for data transmissions to be delayed for the length of time needed to retransmit packets several times. However, in such systems, it may be acceptable for the transmitter  22  to establish a link, and train the receiver  34  before data is ready for transmission. In this case, the transmitter  22  begins by sending a transmission to the receiver to establish a “session”. The transmitter  22  sends special test transmissions, until it receives a response from the receiver  34  indicating the tuning process has completed. 
     The receiver  34  may use a successive approximation method to bring its oscillator frequency close to the oscillator frequency in the transmitter  22 . It is the nature of wireless communications that the BER and RSSI may vary somewhat from transmission to transmission, even with all settings the same. In such a case, the receiver  34  can take average BER or RSSI readings over several transmissions before changing trim settings in circuit  72  or in synthesizer  94 . After this training session is complete, the transmitter  22  is ready to transmit data, without the risk that data packets may be delayed by multiple retransmissions. In this way, it is possible for the transmitter  22  and receiver  34  to achieve improved range/performance compared with a traditional system using even the most accurate (and costly) crystals. 
     In one-way radio systems, there is no ability to use a handshake to cause retransmission of data. In such systems, the “training session” approach may be more suitable. In this case, the transmitter  22  makes a number of packet training transmissions, to which the receiver  34  tunes, before the transmitter  22  begins transmitting “real” data. In broadcast (one-to-many) systems, whether one-way or two-way, the “training session’ approach generally may be more suitable, because of the difficulty of managing handshakes from many receivers. 
     In another implementation, the transmitter  22  or the receiver  34  may intentionally reduce the gain of the transmitted or received signal in order to more quickly and precisely trim the crystal oscillator frequency. For example, during frequency trimming, the distance between the transmitter  22  and receive  34  may be close enough and the signal strength strong enough so that the BER is within some acceptable range, and the BER is so low that it is difficult for the receiver  34  to accurately trim the crystal oscillator frequency. In other words, the transmitter and receiver are so close, the effects of reference frequency offset cannot be accurately detected. However, when the transmitter and receiver are moved further apart, the crystal oscillator frequency offset may cause problems with the signal transmission. 
     To overcome this problem, the transmitter  22  in  FIG. 7  may include a variable amplifier  90  that can be turned down during the crystal oscillator frequency training session. The gain of the transmit signal  28  is intentionally reduced so that the receiver  34  can more effectively determine how much, if any, frequency offset exists between the transmitter  22  and receiver  34 . Similarly, the receiver  34  in  FIG. 8  can include a variable amplifier  97  that is intentionally turned down during the training session. After frequency trimming is complete in either the transmitter  22  and/or receive  34 , the amplifiers  90  ( FIG. 7 ) and/or amplifier  97  ( FIG. 8 ) is returned to their normal operating levels. 
     In Frequency Hopping Spread Spectrum (FHSS) systems, the transmitter  22  and receiver  34  may not stay long enough on a single frequency to complete either a training session or multiple retransmissions on a single frequency. However, in FHSS systems where the carrier frequency of each channel is derived from the same base crystal frequency, either the training session method, or the multiple retry method may be applied. The only difference is that successive retries or successive transmissions within the training session will be on different channels. 
     In another embodiment, the transmitter  22  and receiver  34  may include temperature sensors  120  ( FIG. 5 ). For example, the transmitter  22  or receiver  34  in  FIG. 5  may include firmware that is executed on the microcontroller (MCU)  62  that monitors the temperature sensor  120 . In this embodiment, either the transmitter  22  or the receiver  34  may initiate a new training session any time a temperature change of more than a pre-determined amount is detected by temperature sensor  120 . In yet another embodiment, either the transmitter  22  or the receiver  34  may initiate a training session periodically, for example once per day. 
     Reference frequency tuning has many advantages, including enabling the use of low cost, low precision crystals in wireless systems. A further advantage is that transmitters and receivers can tune their oscillators to each other, enabling better receive sensitivity with low cost crystals than what is generally possible even with the most accurate and expensive crystals. 
     It should be appreciated that reference throughout this specification to “one embodiment” or “an embodiment” means that a particular feature, structure or characteristic described in connection with the embodiment is included in at least one embodiment of the present invention. Therefore, it is emphasized and should be appreciated that two or more references to “an embodiment” or “one embodiment” or “an alternative embodiment” in various portions of this specification are not necessarily all referring to the same embodiment. Furthermore, the particular features, structures, or characteristics may be combined as suitable in one or more embodiments of the invention. 
     Similarly, it should be appreciated that in the foregoing description of exemplary embodiments of the invention, various features of the invention are sometimes grouped together in a single embodiment, figure, or description thereof for the purpose of streamlining the disclosure aiding in the understanding of one or more of the various inventive aspects. This method of disclosure, however, is not to be interpreted as reflecting an intention that the claimed invention requires more features than are expressly recited in each claim. Rather, as the following claims reflect, inventive aspects lie in less than all features of a single foregoing disclosed embodiment. Thus, the claims following the detailed description are hereby expressly incorporated into this detailed description, with each claim standing on its own as a separate embodiment of this invention. 
     The system described above can use dedicated processor systems, micro controllers, programmable logic devices, or microprocessors that perform some or all of the operations. Some of the operations described above may be implemented in software and other operations may be implemented in hardware. 
     For the sake of convenience, the operations are described as various interconnected functional blocks or distinct software modules. This is not necessary, however, and there may be cases where these functional blocks or modules are equivalently aggregated into a single logic device, program, or operation with unclear boundaries. In any event, the functional blocks and software modules or features of the flexible interface can be implemented by themselves, or in combination with other operations in either hardware or software. 
     Having described and illustrated the principles of the invention in a preferred embodiment thereof, it should be apparent that the invention may be modified in arrangement and detail without departing from such principles. I claim all modifications and variation coming within the spirit and scope of the following claims.