Abstract:
A control device for a switching converter having an input terminal, an output terminal, a semi-bridge of a first and second transistor coupled between the input terminal and a reference voltage, includes a first circuit for driving the first transistor and a second circuit for driving the second transistor. The converter further includes a bootstrap circuit for powering the first drive circuit. The bootstrap circuit includes a capacitor coupled between a supply voltage and the common terminal of the first and second transistors. The control device acts upon the second transistor to assure the charging of the capacitor.

Description:
1. RELATED APPLICATION  
       [0001]     The present application claims priority of Italian Patent Application No. MI2005A002055 filed Oct. 27, 2005, which is incorporated herein in its entirety by this reference.  
       2. FIELD OF THE INVENTION  
       [0002]     The present invention relates to a device for controlling a switching converter and the related switching converter.  
       3. BACKGROUND OF THE INVENTION  
       [0003]     Switching converters such as the buck converter shown in  FIG. 1  are generally known in the current state of the art. Said converter comprises an MOS transistor  1  having a non-drivable terminal coupled to an input voltage Vin and another non-drivable terminal coupled to the cathode of an asynchronous rectifier diode D 1  having its anode coupled to ground GND; the transistor is driven by a control device  2 . The cathode of the diode D 1  is coupled to a low-pass filter comprising an inductor L and capacitor C from whose ends the converter output voltage Vout is drawn.  
         [0004]     In conditions of operation with the continuous conduction mode (CCM), that is when the current in the inductor L never goes to zero, and with a resistive type of load LOAD, if the transistor  1  has an “on” time Ton and an “off” time Toff, where T=Ton+Toff, it follows that Vout=D*Vin where D is the duty cycle given by D=Ton/T. In conditions of operation with the discontinuous conduction mode (DCM), that is when the current in the coil goes to zero during the switching period, the output voltage Vout is a function of the value of the inductor L, time period T, duty cycle D and input voltage Vin, i.e.  
       Vout   =       2   ⁢   Vin       1   +       (     1   +         8   ⁢   L       R   ⁢           ⁢   T       *     1     D   2           )     2             
 
 where R is the resistive value of the load LOAD. 
 
         [0005]     Another buck converter layout is shown in  FIG. 2 . The converter comprises a first MOS transistor HS having a non-drivable terminal coupled to the input voltage Vin and another non-drivable terminal P coupled to a terminal of the inductor L and a non-drivable terminal of a second MOS transistor LS coupled to ground GND. The other terminal of inductor L is coupled to the capacitor C, having its other terminal coupled to ground GND; the capacitor C is placed in parallel with the load LOAD and a resistive divider comprising a series of two resistors, R 1  and R 2 . A fraction VFB of the output voltage Vout is input to a control device  20 . The transistors HS and LS are switched on in a push-pull mode and as a result there is a lower power dissipation, given that the voltage drop at the ends of the transistor LS is lower than the voltage drop on the diode.  
         [0006]     The control device  20  comprises a first circuit  24  comprising in turn a comparator for comparing the voltage VFB with a reference voltage Vref and means able to effect a pulse width modulation (PWM) in response to said comparison. The control device  20  comprises two drive circuits or drivers  21  and  22  receiving as inputs the PWM signals output by the circuit  24  and able to drive the transistors HS and LS via the signals HSIDE and LSIDE. The driver  22  is powered by a voltage Vccdr whereas driver  21  is powered by a voltage Vcb originating from a bootstrap circuit  23  comprising a capacitor Cboot situated between the node P and the cathode of a diode Dcb having its anode coupled to the voltage Vccdr.  
         [0007]     When the converter is switched on, the node P is grounded GND and the capacitance Cboot is charged to the voltage Vccdr−Vd where Vd is the voltage drop of the diode Dcb. When a pulse arrives from the PWM signal output by circuit  24 , driver  21  starts to charge the gate of the HS transistor, supplying a charge Q drawn from the capacitance Cboot. When the HS transistor is switched on, the node P is brought to the voltage Vin and the voltage Vcb is forcibly brought to the voltage Vin+Vcboot where Vcboot is the voltage at the ends of the capacitor Cboot. In this condition the driver  21  supplies a voltage to the gate of the HS transistor that is sufficient to keep it on. The switching cycle concludes with the switching off of the transistor HS, whose gate is brought to the voltage of the node P. When the transistor LS is switched on, the node P is again brought to ground GND and the capacitance Cboot is thus recharged via the diode Dcb.  
         [0008]     In switching periods where the node P is not brought to ground GND, the capacitor Cboot, non-recharged, tends to become discharged due to leakage currents and the charging of the capacitance Cgs of the transistor HS. In the event of a sequence of consecutive cycles in which this occurs, the capacitor Cboot may become discharged to a point where the voltage is no longer sufficient to enable the HS transistor to be switched on.  
         [0009]     Such a situation occurs when the device operates with a duty cycle D=1 or when the converter is unable to discharge the current of the inductor L and a residual output current remains; in the latter case, when the device is switched on, the voltage to at the node P is equal to the voltage Vout. In either case the voltage Vgs between the source and gate of the HS transistor will not be sufficient to switch it on.  
       SUMMARY OF THE INVENTION  
       [0010]     In view of the current state of the art, the object of the present invention is to provide a control device for a switching converter that overcomes the aforesaid drawback.  
         [0011]     According to the present invention this object is achieved by means of a switching converter control device having an input terminal and output terminal, said converter comprising a semi-bridge of a first and second transistor coupled between the input terminal and a reference voltage, said control device comprising a first circuit suitable for driving said first transistor and a second circuit suitable for driving said second transistor, said converter comprising a bootstrap circuit suitable for powering said first drive circuit, said bootstrap circuit comprising a capacitor coupled between a supply voltage and the terminal shared by said first and second transistors, further characterized in that it comprises means capable of acting upon said second transistor in order to assure the charging of said capacitor. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0012]     The characteristics and advantages of the present invention will become apparent from the following detailed description of the practical embodiments thereof, illustrated as non-restrictive examples in the appended drawings, in which:  
         [0013]      FIG. 1  is a diagram of a known buck converter layout;  
         [0014]      FIG. 2  is a diagram of another buck converter according to the known prior art;  
         [0015]      FIG. 3  is a diagram of an equivalent capacitance Cboot discharge circuit;  
         [0016]      FIG. 4  is a graph illustrating the trend in the maximum value of the number of non-switching cycles in order for the capacitance Cboot not to be discharged;  
         [0017]      FIG. 5  is a graph illustrating a possible time trend of the current IL;  
         [0018]      FIG. 6  is a diagram of a switching converter provided with a control device according to the present invention;  
         [0019]      FIG. 7  is a diagram of a part of the control device of  FIG. 6  according to a first embodiment of the present invention and a variant thereof; and  
         [0020]      FIG. 8  is a diagram of a part of the control device of  FIG. 6  according to a second embodiment of the present invention. 
     
    
     DETAILED DESCRIPTION  
       [0021]     Considering the buck converter layout illustrated in  FIG. 2 , the aforesaid situation in which the capacitor Cboot discharges to a point where the voltage is no longer sufficient to switch on the HS transistor is overcome when the output voltage Vout becomes such as to enable charging of the capacitor Cboot. The discharging of the capacitor Cboot occurs in a different manner in the two previously mentioned cases. Where the duty cycle D=1, no switching takes place and the capacitor Cboot discharges linearly. Where D&lt;1, the capacitor Cboot will discharge more rapidly due to the charge required by the capacitance Cg, i.e. the capacitance seen from the gate terminal of the transistor HS, which substantially coincides with the capacitance Cgs between the gate and source terminals of the transistor HS, as can be seen from  FIG. 3 . At each switching cycle the voltage Vcb drops by an amount DV: the higher the ratio between the capacitances Cg and Cboot, the higher the voltage drop, i.e.:  
         D   ⁢           ⁢   V     =     Vcb   ⁢       Cg     Cg   +   Cboot       .           
 
 Expressing the voltage Vcb at the Nth cycle as a function of the voltage Vcb at the previous cycle it follows that:  
         Vcb   ⁡     (   N   )       =       Vcb   ⁡     (     N   -   1     )       ⁢       Cg     Cg   +   Cboot       .           
 
 If we indicate as Vcb_max the initial voltage versus the capacitance Cboot it follows that  
         Vcb   ⁡     (   N   )       =     Vcb   -     max   *         [     Cboot     Cboot   +   Cg       ]     N     .             
 
 Given a fixed minimum tolerable voltage Vcb we will have an upper limit G versus N; when this limit is exceeded the capacitance Cboot will discharge to below the pre-established minimum value, i.e.  
         N   &lt;       ln   ⁡     (     Vcb_min   Vcb_max     )         ln   ⁡     (     Cboot     Cboot   +   Cg       )           =     G   .         
 
  FIG. 4  shows a graph of the trend in the maximum number of non-switching cycles in order for the capacitance Cboot not to be discharged, where X=Cg/Cboot and ki=Vcb_min/Vcb_max and where Vcb_max=5V and Vcb_min(k0)=4.5V, Vcb_min(k1)=4.25V, Vcb_min(k2)=4.0V, Vcb_min(k3)=3.75V and Vcb_min(k4)=3.5V. 
 
         [0022]     In the event that the aforesaid condition is met, another problem may arise: it may happen that the converter induces a decrease in the current IL that flows through the inductor L. If the decrease in the current IL during the switching on of the transistor LS in the period Tmask_ 1  is greater than the increase during the time periods Ton, the average current will decrease, becoming negative. To avoid this it is necessary to forcibly impose an “off” time Toff after a number of cycles N greater than  
           Tmask_l   Tsw     *     Vout     Vin   -   Vout         =         Tmask_l   Tsw     *     D     1   -   D         =   H         
 
 where Tmask_ 1  is the descent time of the current IL (with IL(t) (Vout/L)*t) and Tsw is the cycle period so that in the period given by N*Tsw−Tmask_ 1  the current IL(t)((Vin-Vout)/L)*t, as may be seen from  FIG. 5 . From the value of N, the value  
         D   ⁢           ⁢   max     &lt;     1   -     Tmask_l     N   *   Tsw             
 
 is obtained, i.e. the maximum value of the duty cycle D. Therefore, the number of switching cycles N in which it is possible to force the HS transistor off without running into the above-described problems will have a maximum given by G and a minimum given by H. Having thus fixed the value Dmax, i.e. the maximum value of the duty cycle D, the minimum value H of the number of cycles N will likewise be fixed. With this minimum value it is possible to dimension the Cboot/Cg ratio so as to obtain the desired voltage Vcb_min. 
 
         [0023]      FIG. 6  shows a control device of a switching converter according to the invention. The converter comprises a first MOS transistor HS having a non-drivable terminal coupled to the input voltage Vin, present at the converter input terminal IN, and another non-drivable terminal P coupled to a terminal of the inductor L and a non-drivable terminal of a second MOS transistor LS coupled to ground GND. The other terminal of the inductor L is coupled to the converter output terminal OUT and to the capacitor C, whose other terminal is coupled to ground GND; the capacitor C is placed in parallel with the load LOAD and a resistive divider comprising a series of two resistors, R 1  and R 2 . A fraction VFB of the output voltage Vout is input to a control device  200 . The transistors HS and LS are switched on in a push-pull mode and this results in a lower power dissipation given that the voltage drop at the ends of the LS transistor is lower than the voltage drop on the diode.  
         [0024]     The control device  200  comprises a first circuit  201  comprising a comparator for comparing the voltage VFB with a reference voltage Vref and means able to effect a pulse width modulation (PWM) in response to said comparison. The control device  200  comprises two drive circuits or drivers  210  and  220  receiving as inputs the signals HS_ON and LS_ON output by the circuit  201  and which are able to drive the transistors HS and LS via the signals HSIDE and LSIDE. The driver  220  is powered by a voltage Vccdr (e.g. 5 Volts) whereas the driver  210  is powered by a voltage Vcb (e.g. 3 Volts) originating from a bootstrap circuit  230  comprising a capacitor Cboot placed between the node P and cathode of a diode Dcb having its anode coupled to the voltage Vccdr.  
         [0025]     The control device  200  comprises means  100  capable of assuring the correct recharging of the capacitor Cboot in all operating conditions. Said means  100  are able to switch on the transistor LS so as to assure correct recharging of the capacitor Cboot in all operating conditions.  
         [0026]     According to a first embodiment of the invention, said means  100  comprise a first k bit counter  101  and a second k bit counter  102 , as may be better seen from  FIG. 7 . In each switching cycle during which the transistor LS is not switched on, the count value of the counter  101  is increased by one unit; therefore the means  100  comprise a clock device  110  set on the cycle frequency which delivers a pulse per cycle to the counter  101  and a device  111  able to re-initialize the counter  101  via a RESET signal in the presence of a pulse LS_ON. If the Count value of the counter  101  reaches the preset threshold value Count_th of the second counter  102  a signal FORZA_ON_LS is generated to force the transistor LS to switch on. This is carried out by means of a comparator  103  able to compare the Count and Count_th values of the counters  101  and  102  and to generate a signal in response to said comparison. The signal LS_ON has a duration equal to a Tmask_ 1 . When the transistor LS is switched on by means of the signal FORZA_LS_ON or even if the switching on takes places before the Count value reaches the Count_th value, the device  111  will transmit a reset signal to the counter  101 .  
         [0027]     The threshold value Count_th corresponds to the maximum number of cycles in which it is admissible for no switching of the transistor LS to take place; said value depends on the leakage current Ileak, the voltage Vcb, the value of the capacitor Cboot, the frequency of the cycles Fsw and the minimum tolerated voltage value Vcb_min. Where the Ileak and Vcb_min values are known it follows that  
         T   ⁢           ⁢   max     =         (     Vcb   -   Vcb_min     )     *   Cboot     Ileak         
 
 where Tmax is the interval of time beyond which, if no transistor HS and LS switching takes place, the capacitor Cboot will be discharged to the minimum voltage Vcb_min. It follows that Count_th=Tmax*Fsw. 
 
         [0028]     According to a variant of the first embodiment, said control device includes means  300  able to modulate the threshold value Count_th. Every time the transistor HS is switched on, the capacitor Cboot loses a charge Qon which will depend on the voltage Vcb, the capacitance Cboot and the capacitance Cg of the transistor HS. The charge Qon to be supplied to the gate of the transistor HS per switch-on operation may be expressed as follows:  
         Qon   ⁡     (   N   )       =     Ggs   *     Cboot     Cboot   +   Cgs       ⁢       Vcb   ⁡     (     N   -   1     )       .           
 
 The discharge time of the leakage current Ileak is Tleak=Qon/Ileak. The means  300  comprise an n bit counter  301  which receives an input signal HS_ON in order to keep a count of the number of switching operations of the transistor HS; said counter  301  increments its value at each switching of the transistor HS and outputs a signal Ncom in relation to the number of switching operations of the transistor HS. The counter  301  also receives an input RESET signal from the means  111  which serves to re-initialize the counter  301  at the instant the counter  101  is re-initialized. A logic circuit  302  is able to provide a quantity Qdown to the counter  102  so as to decrease the Count_th threshold of said quantity Qdown at every switch-on operation of the transistor HS; said quantity Qdown is variable and determined according to the value Tleak. The first time the transistor HS is switched on, Qdown=Tleak, the second time the transistor HS is switched on, Qdown=(½)*Tleak, the third time the transistor HS is switched on, Qdown=(⅓)*Tleak and so on up to the nth time the transistor HS is switched on, when Qdown (1/n)*Tleak. 
 
         [0029]     Accordingly, the first time the transistor HS is switched on the threshold Count_th is decreased by the quantity Tleak, the second time the transistor HS is switched on the threshold Count_th is decreased by a quantity equal to ½ Tleak, the third time the transistor HS is switched on the threshold Count_th is decreased by a quantity equal to ⅓*Tleak and so on up to the n th  time the transistor HS is switched on, when the threshold Count_th is decreased by a quantity equal to 1/n*Tleak.  
         [0030]     Since in the majority of applications Vcb_min is a value higher than 80% of the normal voltage Vboot, one may assume a Tleak that is constant and independent of the voltage Vcb when a switching on of the transistor HS takes place. As a result, according to another variant of the first embodiment of the inventions the means  300  operate in such a way as to reduce the Count_th value by a fixed amount each time the transistor HS is switched on. In such a case the means  300  will not comprise the counter  301 . The counter  102  receives as an input a RESET signal which re-initializes the counter every time the transistor LS is switched on.  
         [0031]     According to a second embodiment of the present invention, the means  100  of the control device  200  are implemented in a different manner than in the first embodiment, as may be seen from  FIG. 8 . Said means  100  comprise means  400  capable of monitoring the charge status of the capacitor Cboot, for example by means of a comparator  401  which compares the voltage between the node Vcb and the node P with a fixed voltage Vth_boot. When the capacitance Cboot discharges to a point causing the voltage at the pin Vcb to fall below the threshold Vth_hoot, the comparator  401  switches, sending a signal to the circuit  201 , which forces the transistor LS to switch on for a length of time Tmask_ 1 . While the transistor LS is switched on, the voltage at the node P is brought to ground for a period of time equal to TMASK_L. In this phase the voltage at the terminals of diode Dcb is such as to allow the passage of current and provide a voltage Vcb=Vccdr−Vd.  
         [0032]     An application that may benefit from said new control device is to be found in the case of a converter designed to drive a motor under continuous voltage. In such a case the motor speed is controlled simply by varying the voltage applied at the motor input pin (there is no control over torque because the current is not controlled directly). To adjust the motor speed, therefore, the output voltage of the regulator is varied. In these applications the duty cycle depends on the motor speed one wishes to obtain and it is often necessary to bring it to 100% in order to obtain the maximum speed. In such cases the bootstrap capacitance would not be recharged and controller performance would be impaired.  
         [0033]     Many PWM controllers limit the maximum duty-cycle to 80-90% to avoid discharging the bootstrap capacitance but this results in less than optimal performance during the charging transients. According to the control device of the present invention, by contrast, the regulator can function at 100% duty cycle without any problems.  
         [0034]     While there have been described above the principles of the present invention in conjunction with specific memory architectures and methods of operation, it is to be clearly understood that the foregoing description is made only by way of example and not as a limitation to the scope of the invention. Particularly, it is recognized that the teachings of the foregoing disclosure will suggest other modifications to those persons skilled in the relevant art. Such modifications may involve other features which are already known per se and which may be used instead of or in addition to features already described herein. Although claims have been formulated in this application to particular combinations of features, it should be understood that the scope of the disclosure herein also includes any novel feature or any novel combination of features disclosed either explicitly or implicitly or any generalization or modification thereof which would be apparent to persons skilled in the relevant art, whether or not such relates to the same invention as presently claimed in any claim and whether or not it mitigates any or all of the same technical problems as confronted by the present invention. The applicant hereby reserves the right to formulate new claims to such features and/or combinations of such features during the prosecution of the present application or of any further application derived therefrom.