Abstract:
In one aspect, a switch mode power supply comprises a first inductive element; a first switching element, wherein a first end of the first switching element is connected to a second end of the first inductive element; a second inductive element; a second switching element, wherein a first end of the second switching element is connected to a second end of the second inductive element; and wherein a first power source terminal is connected to a first end of the first inductive element and a first end of the second inductive element, and a second power source terminal is connected to a second end of the first switching element and a second end of the second switching element, and wherein a load is connected to the first end of the first switching element and the first end of the second switching element.

Description:
RELATED APPLICATIONS 
   The present application is based on and claims priority from U.S. Provisional Application Ser. No. 60/607,962, filed Sep. 8, 2004, which is hereby incorporated fully by reference. 

   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention is generally in the field of electrical power systems. More specifically, the present invention is in the field of switch mode power supplies. 
   2. Background Art 
   Today, power supplies are critical components of many industrial and household electronics. A power supply is sometimes called a power converter and the process is called power conversion. The Power Sources Manufacturers Association&#39;s (PSMA) Handbook of Standardized Terminology for the Power Sources Industry defines a power supply as “a device for the conversion of available power of one set of characteristics to another set of characteristics to meet specified requirements.” Power supplies may change the characteristics of the power source voltage and/or current, and include AC to AC, AC to DC, DC to AC and DC to DC power supplies. 
   For example, from personal computers to hairdryers and cell phone chargers, many electronic items need to convert the 120 Volts, 60 Hz, AC power found in a home in the United States or 220 Volts, 50 Hz, AC power found in other countries to adapt to appropriate characteristics required by the electronic equipment. As an example, personal computers typically require a 5 Volts DC power to operate their internal electronic components, which is provided by a power supply within personal computer that receives and converts the 120 Volts, AC power to 5 Volts DC power. 
   A switch mode power supply (SMPS) is a power supply that utilizes switches or transistors in conjunction with low loss components such as capacitors, inductors, and transformers. SMPS is known for its low power dissipation, which equates to high efficiency. SMPS has been used for many years in industrial and aerospace applications. Today, SMPS is used extensively in AC powered electronic devices, such as computers, monitors, television receivers, VCRs, etc. 
   A variety of converter topologies are used in SMPS to regulate power. Today, there are a variety of basic topologies in use, including flyback converters, boost converters, single transistor forward converters, half bridge forward converters, full bridge ZVT converters, etc. As an example,  FIG. 1  illustrates conventional half bridge converter  100  along with example related waveforms  130 . As shown, half bridge converter  100  includes switching transistor Q 1   102 , switching transistor Q 2   104 , capacitor C 1  In  106 , capacitor C 2  In  108 , flux balance capacitor  110 , half bridge transformer  112 , diode CR 1   114 , diode CR 2   116 , recirculating diode  118 , capacitor C Out  120 , and output filter inductor  122 . Voltage signals +VIn  124 , −VIn  126 , +VOut  128 , and −VOut  130  are also indicated. In a first stage, when switching transistor Q 1   102  turns on, the voltage is reflected across the output windings, and rectified by diode CR 1   114 , charging output filter inductor  122 . When transistor Q 1   102  turns off, the voltage drive across half bridge transformer  112  primary drops to zero, and energy stored in the leakage and magnetizing inductances causes a turn-off overshoot, which is clamped by the body diode of transistor Q 2   104 . In a second stage, switching transistor Q 2   104  turns on, and half bridge transformer  112  is driven in the opposite direction, resetting the flux balance in the half bridge transformer  112  core. The output of half bridge transformer  112  is connected to a half wave rectifier, so the alternating polarity pulse train is rectified into a unidirectional pulse train of twice the frequency. Output capacitor C Out  120  and output filter inductor  122  store energy and integrate the duty cycle so that the output voltage is proportional to the product of the rectified output voltage and duty cycle. 
   Furthermore, SMPS may use pulse width modulation (PWM) or pulse rate modulation (PRM) to regulate the power. For example, television receivers and computer monitors may use either PWM or PRM, while VCRs typically use PRM supplies. PWM SMPS performs its function by varying the “on” or conduction time of the switches or transistors, such that the frequency of the input signal remains constant while the duty cycle varies. As the width of the pulse is increased, the switching transistor stays on longer, and more energy is applied to the switching transformer, which increases the output voltage. Likewise, as the pulse width is made narrower, the transistor is on for a shorter amount of time, and less energy is applied to the transformer. On the other hand, PRM SMPS varies the rate or frequency at which the switching transistor is turned off and on. As the pulse rate increases, the “on” time of the switch decreases. When the switching transistor is turned on and off at a faster rate, less energy is applied to the transformer. 
   In view of the present state of the art, there remains a strong need for an SMPS topology that can offer a unique control strategy to enable multi-function switches, provide a direct pulse-by-pulse conversion of pulse-width into amplitude, eliminate the dead time between the PWM pulses, eliminate the need for overly complex circuitry, generate less high frequency ripple at the outputs, and improve power density and efficiency. 
   SUMMARY OF THE INVENTION 
   The present invention is directed to direct amplitude modulation for switch mode power supplies (“SMPS”). The invention overcomes the need in the art for an SMPS topology that can offer a unique control strategy to enable multi-function switches, provide a direct pulse-by-pulse conversion of pulse-width into amplitude, eliminate the dead time between the PWM pulses, eliminate the need for overly complex circuitry, generate less high frequency ripple at the outputs, as well as input, improve power density and efficiency, and other advantages. 
   In one aspect of the present invention, a switch mode power supply comprises a first inductive element having a first end and a second end; a first switching element having a first end and a second end, wherein the first end of the first switching element is connected to the second end of the first inductive element; a second switching element having a first end and a second end, wherein the first end of the second switching element is connected to the second end of the first switching element; a second inductive element having a first end and a second end; a third switching element having a first end and a second end; and a fourth switching element having a first end and a second end, wherein the first end of the fourth switching element is connected to the second end of the second inductive element, and wherein the second end of the fourth switching element is connected to the first end of the third switching element; wherein a first power source terminal is connected to the first end of the first inductive element and the first end of the second inductive element, and a second power source terminal is connected to the second end of the second switching element and the second end of the third switching element, and wherein a load is connected to the second end of the first switching element and the second end of the fourth switching element. 
   In an additional aspect, the switch mode power supply further comprises a controller circuit, wherein the control circuit provides a switching signal to each of the first switching element, the second switching element, the third switching element and the fourth switching element. 
   In another aspect, the switch mode power supply operates in four consecutive time periods, including a first time period, a second time period, a third time period and a fourth time period. During the first time period the first switching element is open, the second switching element is closed, the third switching element is open and the fourth switching element is closed. During the second time period the first switching element is closed, the second switching element is closed, the third switching element is open and the fourth switching element is closed. During the third time period the first switching element is closed, the second switching element is open, the third switching element is closed and the fourth switching element is open. During the fourth time period the first switching element is closed, the second switching element is open, the third switching element is closed and the fourth switching element is closed. 
   In another aspect, the switch mode power supply includes additional circuitry for transferring an excess energy in each of the first inductive element and the second inductive element to the load. In yet a different aspect, the switch mode power supply includes additional circuitry for transferring an excess energy in each of the first inductive element and the second inductive element to the power source. 
   In a separate aspect of the present invention, a method of supplying power is provided, which uses a power supply, a first inductive element, a second inductive element, a load, a first switching element, a second switching element, a third switching element and a fourth switching element. The method comprises opening the first switching element, closing the second switching element, opening the third switching element and closing the fourth switching element during a first time period to create a first circuit including the second switching element connected to the load connected to the fourth switching element connected to the power supply connected to the second switching element; closing the first switching element, closing the second switching element, opening the third switching element and closing the fourth switching element during a second time period to create the first circuit, and a second circuit including the second switching element connected to the first switching element connected to the power supply connected to the second switching element; closing the first switching element, opening the second switching element, closing the third switching element and opening the fourth switching element during a third time period to create a third circuit including the third switching element connected to the load connected to the first switching element connected to the power supply connected to the third switching element; and closing the first switching element, opening the second switching element, closing the third switching element and closing the fourth switching element during a fourth time period to create the third circuit, and a fourth circuit including the third switching element connected to the fourth switching element connected to the power supply connected to the third switching element. 
   In another aspect, the method further comprises storing energy in the first inductive element during the second time period; and transferring the energy in the first inductive element to the load during the third time period. Yet, in a different aspect, the method further comprises storing, energy in the second inductive element during the fourth time period; and transferring the energy in the second inductive element to the load during the first time period. 
   In other aspects, the method further comprises transferring an excess energy in each of the first inductive element and the second inductive element to the power source, or transferring an excess energy in each of the first inductive element and the second inductive element to the load. 
   In another separate aspect, a switch mode power supply comprises a first inductive element having a first end and a second end; a first switching element having a first end and a second end, wherein the first end of the first switching element is connected to the second end of the first inductive element; a second inductive element having a first end and a second end; a second switching element having a first end and a second end, wherein the first end of the second switching element is connected to the second end of the second inductive element; and wherein a first power source terminal is connected to the first end of the first inductive element and the first end of the second inductive element, and a second power source terminal is connected to the second end of the first switching element and the second end of the second switching element, and wherein a load is connected to the first end of the first switching element and the first end of the second switching element. 
   Other features and advantages of the present invention will become more readily apparent to those of ordinary skill in the art after reviewing the following detailed description and accompanying drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  illustrates a conventional half bridge converter topology for an SMPS. 
       FIG. 2  illustrates a converter topology for an SMPS, according to one embodiment of the present invention. 
       FIG. 3  illustrates an example timing diagram for the converter of  FIG. 2 . 
       FIG. 4  illustrates two circuits formed in the converter of  FIG. 2  during t 1  and t 2  of  FIG. 3 . 
       FIG. 5  illustrates two circuits formed in the converter of  FIG. 2  during t 3  and t 4  of  FIG. 3 . 
       FIG. 6  illustrates time-to-amplitude conversion for the converter of  FIG. 2  based on the timing diagram of  FIG. 3 . 
       FIG. 7  illustrates a converter for transfer of unused inductive energy to the source, according to one embodiment of present invention. 
       FIG. 8  illustrates a converter for transfer of unused inductive energy to the load, according to one embodiment of present invention. 
       FIG. 9  illustrates an output voltage dependency on the duty cycle for the converter of  FIG. 8 , including three modes of operation PWM, discontinuous and continuous. 
       FIG. 10  illustrates an output waveform at various duty cycles for the converter of  FIG. 8 . 
       FIG. 11  illustrates input current and inductor currents for the converter of  FIG. 8 . 
       FIG. 12  illustrates a converter topology for an SMPS, according to one embodiment of the present invention, depicting two circuits formed in the converter during a first time period. 
       FIG. 13  illustrates the converter topology of  FIG. 12  depicting two circuits formed in the converter during a second time period. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   Although the invention is described with respect to specific embodiments, the principles of the invention can obviously be applied beyond the specifically described embodiments of the invention described herein. Moreover, in the description of the present invention, certain details have been left out to not obscure the inventive aspects of the invention. The details left out are within the knowledge of a person of ordinary skill in the art. The drawings in the present application and their accompanying detailed description are directed to merely example embodiments of the invention. Like or corresponding reference numerals may indicate like or corresponding elements among the figures. To maintain brevity, other embodiments of the invention which use the principles of the present invention are not specifically described in the present application and are not specifically illustrated by the present drawings. 
   Embodiments according to the present invention relate to direct conversion of duty cycle into amplitude in a switch mode power supply or SMPS. As shown herein, an H-bridge converter along with a unique control strategy enables multi-function of the individual switches within the H-bridge converter and direct pulse-by-pulse conversion of pulse-width into amplitude. 
     FIG. 2  illustrates an example converter topology for an SMPS, according to one embodiment of the present invention. H-bridge converter  200  can be used to convert a voltage of a signal, such as a direct current (“DC”) signal, from a first voltage to a second voltage. For example, H-bridge converter  200  can be used to convert a 120V signal to a 240V (step-up) signal in one illustrative embodiment. In another illustrative embodiment, H-bridge converter  200  can be used to convert a 120V signal to a 5V signal (step-down). 
   H-bridge converter  200  comprises source U 1   202 , inductive elements, such as inductor L 1   204 , inductor L 2   206 , load R 1   208 , and switching elements, such as switch S 1   210 , switch S 2   212 , switch S 3   216 , and switch S 4   214 . Control and driver circuitry (not shown) can be coupled to switches S 1   210 , S 2   212 , S 3   216  and S 4   214  to control “on” time and “off” time of the switches. Switches S 1   210 , S 2   212 , S 3   216  and S 4   214  are responsive to switching signals generated by the control circuitry. In one embodiment, switches S 1   210 , S 2   212 , S 3   216  and S 4   214  can be any other suitable transistor, switching devices, IGBT, GTO, MOSFET or any other semiconductor or not-semiconductor based switch. Source  202  is a power source, such as a DC current source (or DC voltage source), in one embodiment. As shown, H-bridge converter  200  comprises two vertical branches, with inductor L 1   204  in one vertical branch and inductor L 2   206  in the other vertical branch. 
   Conventional H-bridge converter allow only two switches, one in each vertical branch to be ON (closed or conducting) at a time, and in order to provide an energy transfer, the two switches are diagonal to each other. In contrast, various embodiments of the present invention do not have such limitations and can allow three and even all four of switches S 1   210 , S 2   212 , S 3   216  and S 4   214  to be ON at a time. For example, as shown below, turning ON three of switches S 1   210 , S 2   212 , S 3   216  and S 4   214  at a time creates two independent electrical circuits, enabling performance of two conversion cycles simultaneously. 
     FIG. 3  depicts example timing diagram  300  for controlling switches S 1   210 , S 2   212 , S 3   216  and S 4   214  by the control circuitry, where the value “1” represents a particular switch being ON (or closed) and the value “0” represents a particular switch being OFF (or open). Accordingly, S 1  timing  310 , S 2  timing  312 , S 3  timing  316  and S 4  timing  314  indicate the ON and OFF states of switches S 1   210 , S 2   212 , S 3   216 , and S 4   214 , respectively. V LOAD  timing  308  corresponds to the voltage across the load, namely the voltage across load R 1   208 . In this illustrative example, an effective duty-cycle (“D”) is chosen to be 80 percent. In other words, the control circuitry controls the switches such that switch S 1   210  is ON 80 percent of the time. The timing diagram of  FIG. 3  shows one cycle of operation divided into four time periods t 1 , t 2 , t 3 , and t 4 . A new cycle begins at the second t 1  time period. 
   As shown in  FIG. 3 , when time period “t 1 ” begins and during this period, switches S 2   212  and S 4   214  are ON, and switches S 1   210  and S 3   216  are OFF. Accordingly, referring to  FIG. 4 , outside dashed line  403  in H-bridge converter  400  defines the current path for time period “t 1 ”. Consequently, energy is transferred from source U 1   402  through inductor L 2   406  (using energy previously stored in inductor L 2   406 ) and switch S 4   414  to load R 1   408  in the negative direction with respect to load R 1   408 , and the circuit is closed through switch S 2   412 . 
   Next, when time period “t 2 ” begins, and during this period, switches S 2   412  and S 4   414  remain ON and, additionally, switch S 1   410  is turned ON, and switch S 3   416  remains OFF. Thus, an additional circuit is created, which is shown by inside dotted line  405  in  FIG. 4 , where this additional circuit enables inductor L 1   404  to store energy for the next half cycle. 
   Now referring to  FIGs. 3 and 5 , when time period “t 3 ” begins and during this period, switch S 1   510  remains ON, switches S 2   512  and  54   514  are turned OFF, and switch S 3   516  is turned ON. As a consequence, the circuit indicated by inside dashed line  507  is created and the energy is transferred from source U 1   502 . Additionally, the energy that was previously stored in inductor L 1   404  (now,  504 ) during time period “t 2 ” is added and applied in the positive direction to load R 1   508 . 
   When time period “t 4 ” begins and during this period, switch S 1   510  remains ON, switch S 2   512  remains OFF, switch S 3   516  remains ON, and switch S 4   514  is turned ON. While the energy transfer continues for the circuit indicated by inside dashed line  507 , a new circuit indicated by outside dotted line  509  is formed. As a result, in preparation for the next quarter-cycle, i.e. next “t 1 ”, inductor L 2   506  stores energy during time period “t 4 ”. At the end of time period “t 4 ,” the cycle is complete and time period “t 1 ” begins once again. As shown in  FIG. 3 , V LOAD    308  is negative for time periods “t 1 ” and “t 2 ”, and positive for time periods “t 3 ” and “t 4 ”. It should be noted that V LOAD    308  could also be rectified if desired, or further stepped up or down by transformer. 
   Table 1 below illustrates various states of H-bridge converter  200  along with a basic set of related equations. 
   
     
       
             
           
         
             
               TABLE 1 
             
             
                 
             
           
           
             
               
                 
                           
                   
                       
                       
                   
                 
               
             
             
                 
             
           
        
       
     
   
   One of ordinary skill in the art can derive from the above equations that, according to one embodiment of the present invention, the output voltage Vo across R 1   408  (or V LOAD ) divided by the input voltage Vin across source U 1   402  is twice the duty cycle or 2D. 
   Advantageously, as described herein, during time period “t 2 ” L 1   404  stores energy that is used during time period “t 3  and t 4 ” to boost the output, and during time period “t 4 ” L 2   506  stores energy that is used during next time period “t 1  and t 2 ” to boost the output. Further, with reference to  FIG. 6 , each of time periods “t 2 ” and “t 4 ” respectively translates directly, pulse-by-pulse, into a voltage in addition to Vin within the next half-cycle. S 1  timing  610 , S 2  timing  612 , S 3  timing  616 , S 4  timing  614  and V LOAD    608  correspond to S 1  timing  310 , S 2  timing  312 , S 3  timing  316 , S 4  timing  314  and V LOAD    308 , respectively. As shown in  FIG. 6 , the duration of HIGH pulse  611  (time period “t 2 ”) determines the magnitude of positive half-period  613 . Similarly, the duration of HIGH pulse  615  (time period “t 4 ”) determines the magnitude of negative half-period  617 . In addition, in embodiments according to the present invention, there is no dead time between the PWM pulses of V LOAD    608 , as shown in  FIG. 6 . 
   Although V LOAD    608  is symmetrical AC, which could be applied to a transformer if desired, in other embodiments, V LOAD  could be made asymmetrical, which can be utilized in special applications, such as a ringing generator with a DC offset. Furthermore, V LOAD  can be rectified and filtered, using various techniques known in the art. 
   In one embodiment of the present invention, the control circuitry operates H-bridge converter  200  in a discontinuous mode. In the discontinuous mode of operation, the energy from each inductor is fully exhausted before switch S 1   210  or switch S 2   212  is turned OFF, respectively. It should be noted that the discontinuous operation of inductors L 1   204  and L 2   206  does not necessarily mean that a discontinuous current is generated at the output of H-bridge converter  200 . This is due to the fact that the load current comprises of interleaved current from both vertical branches. 
   Turning to  FIG. 7 , it illustrates H-bridge converter  700  capable of transferring unused inductive energy to source  702 , according to one embodiment of present invention. The embodiment of  FIG. 7  allows for operation of H-bridge converter  700  in a continuous mode. More specifically, as a result of changes to H-bridge converter  200  of  FIG. 2  to arrive at H-bridge converter  700  of  FIG. 7 , the excess energy stored in air gap of the transformer-inductor L 1   704  and L 2   706  is return to source  702  through the secondary winding of each inductor L 1   704   b  and L 2   706   b  and respective diodes D 1   722  and D 2   724 . It should also be noted that a transformer configuration could also be implemented between the output of the bridge and diodes D 1   722  and D 2   724  that can serve as a rectifier. 
   As shown in  FIG. 7 , H-bridge converter  700  comprises source  702 , transformer-inductor L 1  primary winding  704   a  and secondary winding  704   b , transformer-inductor L 2  primary winding  706   a  and secondary winding  706   b , load R 1   708 , switch S 1   710 , switch S 2   712 , switch S 3   716 , switch S 4   714 , diode D 1   722 , and diode D 2   724 . The timings for switches S 1   710 , S 2   712 , S 3   716  and S 4   714  are similar to the timings for switches S 1   210 , S 2   212 , S 3   216  and S 4   214 , such that H-bridge converter  700  operates similar to H-bridge converter  200 . However, transformer-inductor L 1  secondary winding  704   b  and diode D 1   722  are provided to return previously stored and unused energy in L 1  to source U 1   702 . Similarly, transformer-inductor L 2  secondary winding  706   b  and diode D 2   724  are provided to return unused previously stored energy in L 2  to source U 1   702 . As understood by one of ordinary skill in the art, diodes D 1   722  and D 2   724  cause the current to pass in a single direction from anode of each diode to its cathode, i.e. the direction of source U 1   702 , which facilitates the transfer of excess energy from the inductors to source U 1   702 . 
   Referring now to  FIG. 8 , it illustrates modification to H-bridge converter  200 , such that excess energy in L 1   804  and L 2   806  can be transferred to the load R 1   808 , rather than being transferred to source U 1   702  as shown in  FIG. 7 .  FIG. 8  illustrates H-bridge converter  800  for transfer of unused or excess energy in transformer-inductor L 1  primary winding  804   a  and transformer-inductor L 2  primary winding  806   a  to source U 1   802 , according to one embodiment of present invention. As shown, H-bridge converter  800  comprises source U 1   802 , transformer-inductor L 1  primary winding  804   a  and secondary winding  804   b , transformer-inductor L 2  primary winding  806   a  and secondary winding  806   b , load R 1   808 , switch S 1   810 , switch S 2   812 , switch S 3   816 , switch S 4   814 , diode D 1   823 , diode D 2   825 , diode D 3   826 , diode D 4   828 , diode D 5   822 , diode D 6   824 , and capacitor C 1   830 . It is noted that load R 1   808  and capacitor C 1   830  are coupled in parallel. Capacitor C 1  serves the purpose of an output filter for load R 1 . 
   The operation of H-bridge converter  800  is somewhat analogous to the operation of H-bridge converter  700 . However, the excess energy in transformer-inductor L 1  primary winding  804   a  and transformer-inductor L 2  primary winding  806   a  is transferred to the load in H-bridge converter  800 , as opposed to the source in H-bridge converter  700 . Current or energy can pass through transformer-inductor L 1  secondary winding  804   b  through diode D 5   822  to node  831 . Current can also pass through transformer-inductor L 2  secondary winding  806   b  through diode D 6   824  to node  831 . Current can further pass through diodes D 1   823  through D 4   828  to node  831 . From node  831 , energy or current can pass to load R 1   808 , or also through switches S 4   814  and S 3   816 . As understood by one of ordinary skill in the art, diodes in  FIG. 8  cause the current to pass in a single direction, i.e. the direction of load R 1   808 , which facilitates the transfer of excess energy in the inductors to the load. 
   The specification and components details of an experimental H-bridge converter  800  are shown below in Table 2. 
   
     
       
             
           
             
             
             
             
           
             
           
             
             
             
           
         
             
               TABLE 2 
             
             
                 
             
           
           
             
               EXPERIMENTAL CONVERTER 
             
             
               SPECIFICATION 
             
             
                 
             
           
        
         
             
                 
               FSW 
               [kHz] 
               100 
             
             
                 
               Vin 
               [V] 
               48 
             
             
                 
               Vo 
               [V] 
               48 
             
             
                 
               PoMAX 
               [W] 
               750 
             
             
                 
               ΔVoMAX 
               [V] 
               0.5 
             
             
                 
               Δlin MAX 
               [A] 
               5 
             
             
                 
               EFF 
               [%] 
               92 
             
             
                 
                 
             
           
        
         
             
               COMPONENTS 
             
             
                 
             
           
        
         
             
                 
               S1 through S4 
               IRF3710ZS 
             
             
                 
               L1, L2 
               20 uH/18 A 
             
             
                 
               D1 through D4 
               100BGQ100J 
             
             
                 
               C1 
               200 uF/100 V 
             
             
                 
               U1, U2 (drivers) 
               IR2085S 
             
             
                 
               Arbitrary generator 
               AWG400 
             
             
                 
                 
             
           
        
       
     
   
   Referring to  FIG. 9 , graph  900  illustrates an output voltage dependency on the duty cycle for the converter of  FIG. 8 , including three modes of operation PWM, discontinuous and continuous. Vertical axis  902  represents output voltage (Vo) in Volts and horizontal axis  904  represents the effective duty cycle (D) in percentage. Three different modes of operation are shown in  FIG. 9  to be PWM mode  906 , discontinuous mode  908 , and continuous mode  910 . Experimental output voltage dependency indicator  912  and simulated output voltage dependency indicator  914  are shown, spanning the three modes of operation. As shown, below a duty cycle of approximately fifty percent (50%), H-bridge converter  800  is operated in PWM mode  906  and basically functions as a regular bridge. The amplitude of the PWM pulses does not change with the pulse width (i.e. with the duty cycle). For a duty cycle ranging from approximately fifty percent (50%) to approximately ninety percent (90%), H-bridge converter  800  operates in an enhanced mode in discontinuous mode  908 . For a duty cycle greater than approximately ninety percent (90%), H-bridge converter  800  operates in continuous mode  910 . 
   Referring to  FIG. 10 , graph  1000  illustrates an output waveform at various duty cycles for the converter of  FIG. 8 . Specifically, Vb in is plotted in Volts on vertical axis  1002 . Each division in the vertical direction is equal to 20 volts. Horizontal axis  1004  represents time. As indicated, buck-boost boundary  1006  lies at a duty cycle of approximately seventy percent (70%); in this particular example, a voltage of approximately 48 volts. For a duty cycle above approximately fifty percent (50%), the dead time is not present and the amplitude of the PWM pulses is changing with the effective duty cycle. After passing a duty cycle of about ninety percent (90%), the inductors run in continuous mode. It is important to note that while the current within individual inductors may be discontinuous, the output current may still—due to the fact that both vertical branches are interleaved—be continuous. 
     FIG. 11  illustrates graph  1100  depicting input current and inductor currents for H-bridge converter  800  of  FIG. 8 . For example, corresponding inductors and input current waveforms for a duty cycle of fifty percent (50%) (discontinuous mode) and a duty cycle of one-hundred percent (100%) (continuous mode) are presented. When the duty cycle is one-hundred percent (100%), switches S 1   810  and S 4   814  are permanently ON and essentially eliminated. Consequently, input ripple is substantially eliminated, because the summation of the inductor currents is constant at any point of time. This configuration will also eliminate the ability to control the output voltage. The waveform may be shifted up and down using asymmetrical control. 
   A duty cycle of one-hundred percent (100%) means that both S 1   210  and S 4   214  switches within the H-bridge converter  200  of  FIG. 2  remain ON at all time, while the other two switches, i.e. S 2   212  and S 3   216 , are switched alternately ON and OFF. This embodiment is depicted in  FIG. 12 , as another embodiment of the present invention.  FIG. 12  illustrates two-switch H-bridge converter  1200 , with switch S 1   1212  ON and switch S 2   1216  OFF, according to one embodiment of the present invention. H-bridge converter  1200  comprises source  1202 , inductor L 1   1204 , inductor L 2   1206 , load R 1   1208 , switch S 1   1212 , and switch S 2   1216 . 
   With reference to  FIG. 12 , in order to achieve symmetrical waveforms, the duty cycle of each switch should be fifty percent (50%), with 180 degrees of offset between the switches. Special applications, for example ringing generators, may require asymmetrical waveforms. This symmetry can be achieved by varying the duty-cycle further. The functionality of H-bridge  1200  depicted in  FIG. 12  is described as follows. When switch S 1   1212  is ON, two circuits are conducting current independently. First circuit  1205  comprises source U 1   1202 , inductor L 1   1204 , and switch S 1   1212 . During this half cycle, energy is stored in inductor L 1   1204 . Simultaneously, second circuit  1203  is created, which provides a current path including source U 1   1202 , inductor L 2   1206 , load R 1   1208 , and switch S 1   1212 . In second circuit  1203 , the energy from source U 1   1202  plus the energy stored in inductor L 2   1206  during the previous half cycle are delivered to load R 1   1208 . 
   Referring to  FIG. 13 , H-bridge converter  1300  comprises source  1302 , transformer L 1   1304 , transformer L 2   1306 , load R 1   1308 , switch S 1   1312 , and switch S 2   1316 . When switch S 1   1312  is OFF and switch S 2   1316  is ON, first circuit  1307  and second circuit  1309  are created. Inductor L 2   1306  will be storing energy, while source U 1   1302  and inductor L 1   1304 , using the energy stored during the previous half cycle, supply current to load R 1   1308 . The polarity across load R 1   1308  will be opposite during this half period, as shown in  FIGS. 13 and 14 . The cycles described in  FIGS. 13 and 14  will continue repeating. 
   From the above description of the invention it is manifest that various techniques can be used for implementing the concepts of the present invention without departing from its scope. Moreover, while the invention has been described with specific reference to certain embodiments, a person of ordinary skill in the art would recognize that changes can be made in form and detail without departing from the spirit and the scope of the invention. The described embodiments are to be considered in all respects as illustrative and not restrictive. It should also be understood that the invention is not limited to the particular embodiments described herein, but is capable of many rearrangements, modifications, and substitutions without departing from the scope of the invention.