Abstract:
Apparatus and methods of ultra-wideband communication are provided. In one embodiment, a communication frame is generated and includes a synchronization section, and at least two data sections and at least two preamble sections. An ultra-wideband receiver synchronizes with the ultra-wideband transmitter using the synchronization section, and may also re-synchronize with the transmitter using at least one preamble section. In another embodiment, the length of the data sections are selected to ensure that drift between the ultra-wideband receiver and transmitter will be minimal before re-synchronization. This Abstract is provided for the sole purpose of complying with the Abstract requirement rules that allow a reader to quickly ascertain the subject matter of the disclosure contained herein. This Abstract is submitted with the explicit understanding that it will not be used to interpret or to limit the scope or the meaning of the claims.

Description:
[0001]     This is a continuation-in-part of co-pending U.S. patent application Ser. No. 10/948,099, filed Sep. 23, 2004, entitled “Systems and Methods for Implementing an Open Loop Architecture in a Wireless Communication Network,” which itself claims priority under 35 U.S.C. Section 119 to U.S. provisional patent application Ser. No. 60/547,221, entitled “Systems and Methods for Implementing an Open Loop Architecture in a Wireless Communication Network,” filed Feb. 23, 2004, which is a continuation-in-part of co-pending U.S. patent application Ser. No. 10/120,456, entitled “Systems and Methods for Recovering Bandwidth in a Wireless Communication Network,” filed Apr. 9, 2002, which is a continuation-in-part of co-pending U.S. patent application Ser. No. 10/010,601, filed Dec. 6, 2001, entitled “Systems and Methods for Wireless Communication over a Wide Bandwidth Channel using a Plurality of Sub-Channels.” 
     
    
     BACKGROUND OF THE INVENTION  
       [0002]     1. Field of the Invention  
         [0003]     The invention relates generally to ultra-wideband communications, and more particularly to systems and methods for communication using ultra-wideband technology.  
         [0004]     2. Background  
         [0005]     Wireless communication systems are proliferating at the Wide Area Network (WAN), Local Area Network (LAN), and Personal Area Network (PAN) levels. These wireless communication systems use a variety of techniques to allow simultaneous access to multiple users. The most common of these techniques are Frequency Division Multiple Access (FDMA), which assigns specific frequencies to each user, Time Division Multiple Access (TDMA), which assigns particular time slots to each user, and Code Division Multiple Access (CDMA), which assigns specific codes to each user. But these wireless communication systems and various modulation techniques are afflicted by a host of problems that limit the capacity and the quality of service provided to the users. The following paragraphs briefly describe a few of these problems for the purpose of illustration.  
         [0006]     One problem that can exist in a wireless communication system is multipath interference. Multipath interference, or multipath, occurs because some of the energy in a transmitted wireless signal bounces off of obstacles, such as buildings or mountains, as it travels from source to destination. The obstacles in effect create reflections of the transmitted signal and the more obstacles there are, the more reflections they generate. The reflections then travel along their own transmission paths to the destination (or receiver). The reflections will contain the same information as the original signal; however, because of the differing transmission path lengths, the reflected signals will be out of phase with the original signal. As a result, they will often combine destructively with the original signal in the receiver. This is referred to as fading. To combat fading, current systems typically try to estimate the multipath effects and then compensate for them in the receiver using an equalizer. In practice, however, it is very difficult to achieve effective multipath compensation.  
         [0007]     A second problem that can affect the operation of wireless communication systems is interference from adjacent communication cells within the system. In FDMA/TDMA systems, this type of interference is prevented through a frequency reuse plan. Under a frequency reuse plan, available communication frequencies are allocated to communication cells within the communication system such that the same frequency will not be used in adjacent ceils. Essentially, the available frequencies are split into groups. The number of groups is termed the reuse factor. Then the communication cells are grouped into clusters, each cluster containing the same number of cells as there are frequency groups. Each frequency group is then assigned to a cell in each cluster. Thus, if a frequency reuse factor of 7 is used, for example, then a particular communication frequency will be used only once in every seven communication cells. As a result, in any group of seven communication cells, each cell can only use {fraction (1/7)} th  of the available frequencies, i.e., each cell is only able to use {fraction (1/7)} th  of the available bandwidth.  
         [0008]     In a CDMA communication system, each cell uses the same wideband communication channel. In order to avoid interference with adjacent cells, each communication cell uses a particular set of spread spectrum codes to differentiate communications within the cell from those originating outside of the cell. Thus, CDMA systems preserve the bandwidth in the sense that they avoid limitations inherent to conventional reuse planning. But as will be discussed, there are other issues that limit the bandwidth in CDMA systems as well.  
         [0009]     Thus, in overcoming interference, system bandwidth is often sacrificed. Bandwidth is becoming a very valuable commodity as wireless communication systems continue to expand by adding more and more users. Therefore, trading off bandwidth for system performance is a costly, albeit necessary, proposition that is inherent in all wireless communication systems.  
         [0010]     The foregoing are just two examples of the types of problems that can affect conventional wireless communication systems. The examples also illustrate that there are many aspects of wireless communication system performance that can be improved through systems and methods that, for example, reduce interference, increase bandwidth, or both.  
         [0011]     Ultra-wideband (UWB) communications systems, while somewhat more resistant to multipath, also suffer from its effects. UWB is a pulsed form of communications wherein the continuous carrier wave of traditional communications is replaced with discrete pulses of electro-magnetic energy. Some UWB communications systems employ modulation techniques where the data is carried by the precise timing of pulses. As described above, reflected energy travels a different path from the transmitter to the receiver. The path length additionally causes the reflected energy to arrive at the receiver at a different time. Since some UWB systems use timing to impart data, reflected copies of pulses may interfere with the demodulation of the UWB signal.  
         [0012]     Not only are conventional wireless communication systems effected by problems, such as those described in the preceding paragraphs, but also different types of systems are effected in different ways and to different degrees. Wireless communication systems can be split into three types: 1) line-of-sight systems, which can include point-to-point or point-to-multipoint systems; 2) indoor non-line of sight systems; and 3) outdoor systems such as wireless WANs. Line-of-sight systems are least affected by the problems described above, while indoor systems are more affected, due for example to signals bouncing off of building walls. Outdoor systems are by far the most affected of the three systems. Because these types of problems are limiting factors in the design of wireless transmitters and receivers, such designs must be tailored to the specific types of system in which it will operate. In practice, each type of system implements unique communication standards that address the issues unique to the particular type of system. Even if an indoor system used the same communication protocols and modulation techniques as an outdoor system, for example, the receiver designs would still be different because multipath and other problems are unique to a given type of system and must be addressed with unique solutions. This would not necessarily be the case if cost efficient and effective methodologies can be developed to combat such problems as described above that build in programmability so that a device can be reconfigured for different types of systems and still maintain superior performance.  
       SUMMARY  
       [0013]     In order to combat the above problems, apparatus and methods described herein provide novel ultra-wideband communication apparatus, systems and methods. In one embodiment of the present invention, a communication frame is generated and includes a synchronization section, and at least two data sections and at least two preamble sections. An ultra-wideband receiver synchronizes with the ultra-wideband transmitter using the synchronization section, and may also re-synchronize with the transmitter using at least one preamble section. In another embodiment, the length of the data sections are selected to ensure that drift between the ultra-wideband receiver and transmitter will be minimal before re-synchronization.  
         [0014]     These and other features and advantages of the present invention will be appreciated from review of the following Detailed Description of the Preferred Embodiments, along with the accompanying figures in which like reference numerals are used to describe the same, similar or corresponding parts in the several views of the drawings. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0015]     Preferred embodiments of the present inventions taught herein are illustrated by way of example, and not by way of limitation, in the figures of the accompanying drawings, in which:  
         [0016]      FIG. 1  is a diagram illustrating an example embodiment of a wideband channel divided into a plurality of sub-channels in accordance with the invention;  
         [0017]      FIG. 2  is a diagram illustrating the effects of multipath in a wireless communication system;  
         [0018]      FIG. 3  is a diagram illustrating another example embodiment of a wideband communication channel divided into a plurality of sub-channels in accordance with the invention;  
         [0019]      FIG. 4  is a diagram illustrating the application of a roll-off factor to the sub-channels of  FIGS. 1, 2  and  3 ;  
         [0020]      FIG. 5A  is a diagram illustrating the assignment of sub-channels for a wideband communication channel in accordance with the invention;  
         [0021]      FIG. 5B  is a diagram illustrating the assignment of time slots for a wideband communication channel in accordance with the invention;  
         [0022]      FIG. 6  is a diagram illustrating an example embodiment of a wireless communication in accordance with one embodiment of the invention;  
         [0023]      FIG. 7  is a diagram illustrating the use of synchronization codes in the wireless communication system of  FIG. 6  in accordance with the invention;  
         [0024]      FIG. 8  is a diagram illustrating a correlator that can be used to correlate synchronization codes in the wireless communication system of  FIG. 6 ;  
         [0025]      FIG. 9  is a diagram illustrating synchronization code correlation in accordance with the invention;  
         [0026]      FIG. 10  is a diagram illustrating the cross-correlation properties of synchronization codes configured in accordance with the invention;  
         [0027]      FIG. 11  is a diagram illustrating another example embodiment of a wireless communication system in accordance with the invention;  
         [0028]      FIG. 12A  is a diagram illustrating how sub-channels of a wideband communication channel according to the present invention can be grouped in accordance with the present invention;  
         [0029]      FIG. 12B  is a diagram illustrating the assignment of the groups of sub-channels of  FIG. 12A  in accordance with the invention;  
         [0030]      FIG. 13  is a diagram illustrating the group assignments of  FIG. 12B  in the time domain;  
         [0031]      FIG. 14  is a flow chart illustrating the assignment of sub-channels based on SIR measurements in the wireless communication system of  FIG. 11  in accordance with the invention;  
         [0032]      FIG. 15  is a logical block diagram of an example embodiment of transmitter configured in accordance with the invention;  
         [0033]      FIG. 16  is a logical block diagram of an example embodiment of a modulator configured in accordance with the present invention for use in the transmitter of  FIG. 15 ;  
         [0034]      FIG. 17  is a diagram illustrating an example embodiment of a rate controller configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0035]      FIG. 18  is a diagram illustrating another example embodiment of a rate controller configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0036]      FIG. 19  is a diagram illustrating an example embodiment of a frequency encoder configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0037]      FIG. 20  is a logical block diagram of an example embodiment of a TDM/FDM block configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0038]      FIG. 21  is a logical block diagram of another example embodiment of a TDM/FDM block configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0039]      FIG. 22  is a logical block diagram of an example embodiment of a frequency shifter configured in accordance with the invention for use in the modulator of  FIG. 16 ;  
         [0040]      FIG. 23  is a logical block diagram of a receiver configured in accordance with the invention;  
         [0041]      FIG. 24  is a logical block diagram of an example embodiment of a demodulator configured in accordance with the invention for use in the receiver of  FIG. 23 ;  
         [0042]      FIG. 25  is a logical block diagram of an example embodiment of an equalizer configured in accordance with the present invention for use in the demodulator of  FIG. 24 ;  
         [0043]      FIG. 26  is a logical block diagram of an example embodiment of a wireless communication device configured in accordance with the invention:  
         [0044]      FIG. 27  is a flow chart illustrating an exemplary method for recovering bandwidth in a wireless communication network in accordance with the invention;  
         [0045]      FIG. 28  is a diagram illustrating an exemplary wireless communication network in which the method of  FIG. 27  can be implemented;  
         [0046]      FIG. 29  is a logical block diagram illustrating an exemplary transmitter that can be used in the network of  FIG. 28  to implement the method of  FIG. 27 ;  
         [0047]      FIG. 30  is a logical block diagram illustrating another exemplary transmitter that can be used in the network of  FIG. 28  to implement the method of  FIG. 27 ;  
         [0048]      FIG. 31  is a diagram illustrating another exemplary wireless communication network in which the method of  FIG. 27  can be implemented;  
         [0049]      FIG. 32  is a diagram illustrating an example wireless communication system;  
         [0050]      FIG. 33  is a diagram illustrating sampling of a signal received by a receive in the system of  FIG. 32 ;  
         [0051]      FIG. 34  is a diagram illustrating a frame structure that can be used to implement an open loop architecture in the system of  FIG. 32  in accordance with one embodiment;  
         [0052]      FIG. 35  is a diagram illustrating an example embodiment of SYNC packet that can comprise a portion of the frame structure of  FIG. 34 ;  
         [0053]      FIG. 36  is a diagram illustrating correlation of the frame illustrated in  FIG. 34  by a receiver in the system of  FIG. 32 ;  
         [0054]      FIG. 37  is a diagram illustrating OFDM and CDMA communication protocols;  
         [0055]      FIG. 38  is a diagram of a wireless communication system that can be configured to enable coexistence of a plurality of communication protocols, such as the OFDM and CDMA protocols illustrated in  FIG. 37 ;  
         [0056]      FIG. 39  is a diagram of a receiver configured to receive a frame used to achieve coexistence in the system of  FIG. 38 ;  
         [0057]      FIG. 40  is a diagram of another receiver configured to receive a frame used to achieve coexistence in the system of  FIG. 38 ;  
         [0058]      FIG. 41  is a diagram of a wireless communication system configured to enable position location in accordance with one embodiment;  
         [0059]      FIG. 42  is a diagram illustrating example receive circuitry configured to provide the position location within the system of  FIG. 41  in accordance with one embodiment;  
         [0060]      FIG. 43  is an illustration of different communication methods; and  
         [0061]      FIG. 44  is an illustration of two ultra-wideband pulses. 
     
    
       [0062]     It will be recognized that some or all of the Figures are schematic representations for purposes of illustration and do not necessarily depict the actual relative sizes or locations of the elements shown. The Figures are provided for the purpose of illustrating one or more embodiments of the invention with the explicit understanding that they will not be used to limit the scope or the meaning of the claims.  
       DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0063]     In the following paragraphs, the present invention will be described in detail by way of example with reference to the attached drawings. While this invention is capable of embodiment in many different forms, there is shown in the drawings and will herein be described in detail specific embodiments, with the understanding that the present disclosure is to be considered as an example of the principles of the invention and not intended to limit the invention to the specific embodiments shown and described. That is, throughout this description, the embodiments and examples shown should be considered as exemplars, rather than as limitations on the present invention. As used herein, the “present invention” refers to any one of the embodiments of the invention described herein, and any equivalents. Furthermore, reference to various feature(s) of the “present invention” throughout this document does not mean that all claimed embodiments or methods must include the referenced feature(s).  
         [heading-0064]     1. Introduction  
         [0065]     In order to improve wireless communication system performance and allow a single device to move from one type of system to another, while still maintaining superior performance, the systems and methods described herein provide various communication methodologies that enhance performance of transmitters and receivers with regard to various common problems that afflict such systems and that allow the transmitters and/or receivers to be reconfigured for optimal performance in a variety of systems. Accordingly, the systems and methods described herein define a channel access protocol that uses a common wideband communication channel—for all communication cells. The wideband channel, however, is then divided into a plurality of sub-channels. Different sub-channels are then assigned to one or more users within each cell. But the base station, or service access point, within each cell transmits one message that occupies the entire bandwidth of the wideband channel. Each user&#39;s communication device receives the entire message, but only decodes those portions of the message that reside in sub-channels assigned to the user. For a point-to-point system, for example, a single user may be assigned all sub-channels and, therefore, has the full wide band channel available to them. In a wireless WAN, on the other hand, the sub-channels may be divided among a plurality of users.  
         [0066]     In the descriptions of example embodiments that follow, implementation differences, or unique concerns, relating to different types of systems will be pointed out to the extent possible. But it should be understood that the systems and methods described herein are applicable to any type of communication systems. In addition, terms such as communication cell, base station, service access point, etc. are used interchangeably to refer to the common aspects of networks at these different levels.  
         [0067]     To begin illustrating the advantages of the systems and methods described herein, one can start by looking at the multipath effects for a single wideband communication channel  100  of bandwidth B as shown in  FIG. 1 . Communications sent over channel  100  in a traditional wireless communication system will comprise digital data symbols, or symbols, that are encoded and modulated onto a RF carrier that is centered at frequency f c  and occupies bandwidth B. Generally, the width of the symbols (or the symbol duration) T is defined as 1/B. Thus, if the bandwidth B is equal to 100 MHz, then the symbol duration T is defined by the following equation: 
 
 T= 1/ B= {fraction (1/100)} megahertz ( MHZ )=10 nanoseconds ( ns ).  (1) 
 
         [0068]     When a receiver receives the communication, demodulates it, and then decodes it, it will recreate a stream  104  of data symbols  106  as illustrated in  FIG. 2 . But the receiver will also receive multipath versions  108  of the same data stream. Because multipath data streams  108  are delayed in time relative to the data stream  104  by delays d 1 , d 2 , d 3 , and d 4 , for example, they may combine destructively with data stream  104 .  
         [0069]     A delay spread d s  is defined as the delay from reception of data stream  104  to the reception of the last multipath data stream  108  that interferes with the reception of data stream  104 . Thus, in the example illustrated in  FIG. 2 , the delay spread d s  is equal to delay d 4 . The delay spread d s  will vary for different environments. An environment with a lot of obstacles will create a lot of multipath reflections. Thus, the delay spread d s  will be longer. Experiments have shown that for outdoor WAN type environments, the delay spread d s  can be as long as 20 microseconds (20 μs). Using the 10 ns symbol duration of equation (1), this translates to 2000 symbols. Thus, with a very large bandwidth, such as 100 MHz, multipath interference can cause a significant amount of interference at the symbol level for which adequate compensation is difficult to achieve. This is true even for indoor environments. For indoor LAN type systems, the delay spread d s  is significantly shorter, typically about 10 s. For a 10 ns symbol duration, this is equivalent to 100 symbols, which is more manageable but still significant.  
         [0070]     By segmenting the bandwidth B into a plurality of sub-channels  200 , as illustrated in  FIG. 3 , and generating a distinct data stream for each sub-channel, the multipath effect can be reduced to a much more manageable level. For example, if the bandwidth B of each sub-channel  200  is 500 KHz, then the symbol duration is 2 μs. Thus, the delay spread d s  for each sub-channel is equivalent to only 10 symbols (outdoor) or half a symbol (indoor). Thus, by breaking up a message that occupies the entire bandwidth B into discrete messages, each occupying the bandwidth B of sub-channels  200 , a very wideband signal that suffers from relatively minor multipath effects is created.  
         [0071]     Before discussing further features and advantages of using a wideband communication channel segmented into a plurality of sub-channels as described, certain aspects of the sub-channels will be explained in more detail. Referring back to  FIG. 3 , the overall bandwidth B is segmented into N sub-channels center at frequencies f o  to f N−1 . Thus, the sub-channel  200  that is immediately to the right of fc is offset from fc by b/2, where b is the bandwidth of each sub-channel  200 . The next sub-channel  200  is offset by 3b/2, the next by 5b/2, and so on. To the left of fc, each sub-channel  200  is offset by −b/2, −3b/2, −5b/2, etc.  
         [0072]     Preferably, sub-channels  200  are non-overlapping as this allows each sub-channel to be processed independently in the receiver. To accomplish this, a roll-off factor is preferably applied to the signals in each sub-channel in a pulse-shaping step. The effect of such a pulse-shaping step is illustrated in  FIG. 3  by the non-rectangular shape of the pulses in each sub-channel  200 . Thus, the bandwidth b of each sub-channel can be represented by an equation such as the following: 
 
 b =( l+r )/ T;   (2) 
        Where r=the roll-off factor; and     T=the symbol duration.        
 
         [0075]     Without the roll-off factor, i.e., b=1/T, the pulse shape would be rectangular in the frequency domain, which corresponds to a (sin x)/x function in the time domain. The time domain signal for a (sin x)/x signal  400  is shown in  FIG. 4  in order to illustrate the problems associated with a rectangular pulse shape and the need to use a roll-off factor.  
         [0076]     As can be seen, main lobe  402  comprises almost all of signal  400 . But some of the signal also resides in side lobes  404 , which stretch-out indefinitely in both directions from main lobe  402 . Side lobes  404  make processing signal  400  much more difficult, which increases the complexity of the receiver. Applying a roll-off factor r, as in equation (2), causes signal  400  to decay faster, reducing the number of side lobes  404 . Thus, increasing the roll-off factor decreases the length of signal  400 , i.e., signal  400  becomes shorter in time. But including the roll-off factor also decreases the available bandwidth in each sub-channel  200 . Therefore, r must be selected so as to reduce the number of side lobes  404  to a sufficient number, e.g., 15, while still maximizing the available bandwidth in each sub-channel  200 .  
         [0077]     Thus, the overall bandwidth B for communication channel  200  is given by the following equation: 
 
 B=N (1 +r )/ T;   (3) 
        or 
 
 B=M/T;   (4) 
    Where 
 
 M =(1 +r ) N.   (5) 
       
 
         [0080]     For efficiency purposes related to transmitter design, it is preferable that r is chosen so that M in equation (5) is an integer. Choosing r so that M is an integer allows for more efficient transmitters designs using, for example, Inverse Fast Fourier Transform (IFFT) techniques. Since M=N+N(r), and N is always an integer, this means that r must be chosen so that N(r) is an integer. Generally, it is preferable for r to be between 0.1 and 0.5. Therefore, if N is 16, for example, then 0.5 could be selected for r so that N(r) is an integer. Alternatively, if a value for r is chosen in the above example so that N(r) is not an integer, B can be made slightly wider than M/T to compensate. In this case, it is still preferable that r be chosen so that N(r) is approximately an integer.  
         [heading-0081]     2. Example Embodiment of a Wireless Communication System  
         [0082]     With the above in mind,  FIG. 6  illustrates an example communication system  600  comprising a plurality of cells  602  that each use a common wideband communication channel to communicate with communication devices  604  within each cell  602 . The common communication channel is a wideband communication channel as described above. Each communication cell  602  is defined as the coverage area of a base station, or service access point,  606  within the cell. One such base station  606  is shown for illustration in  FIG. 6 . For purposes of this specification and the claims that follow, the term base station will be used generically to refer to a device that provides wireless access to the wireless communication system for a plurality of communication devices, whether the system is a line of sight, indoor, or outdoor system.  
         [0083]     Because each cell  602  uses the same communication channel signals in one cell  602  must be distinguishable from signals in adjacent cells  602 . To differentiate signals from one cell  602  to another, adjacent base stations  606  use different synchronization codes according to a code reuse plan. In  FIG. 6 , system  600  uses a synchronization code reuse factor of 4, although the reuse factor can vary depending on the application.  
         [0084]     Preferably, the synchronization code is periodically inserted into a communication from a base station  606  to a communication device  604  as illustrated in  FIG. 7 , After a predetermined number of data packets  702 , in this case two, the particular synchronization code  704  is inserted into the information being transmitted by each base station  606 . A synchronization code is a sequence of data bits known to both the base station  606  and any communication devices  604  with which it is communicating. The synchronization code allows such a communication device  604  to synchronize its timing to that of base station  606 , which, in turn, allows device  604  to decode the data properly. Thus, in cell  1  (see lightly shaded cells  602  in  FIG. 6 ), for example, synchronization code  1  (SYNC 1 ) is inserted into data stream  706 , which is generated by base station  606  in cell  1 , after every two packets  702 ; in cell  2  SYNC 2  is inserted after every two packets  702 ; in cell  3  SYNC 3  is inserted; and in cell  4  SYNC 4  is inserted. Use of the synchronization codes is discussed in more detail below.  
         [0085]     In  FIG. 5A  an example wideband communication channel  500  for use in communication system  600  is divided into 16 sub-channels  502 , centered at frequencies f 0  to f 15 . A base station  606  at the center of each communication cell  602  transmits a single packet occupying the whole bandwidth B of wideband channel  500 . Such a packet is illustrated by packet  504  in  FIG. 5B . Packet  504  comprises sub-packets  506  that are encoded with a frequency offset corresponding to one of sub-channels  502 . Sub-packets  506  in effect define available time slots in packet  504 . Similarly, sub-channels  502  can be said to define available frequency bins in communication channel  500 . Therefore, the resources available in communication cell  602  are time slots  506  and frequency bins  502 , which can be assigned to different communication devices  604  within each cell  602 .  
         [0086]     Thus, for example, frequency bins  502  and time slots  506  can be assigned to 4 different communication devices  604  within a cell  602  as shown in  FIG. 5 , Each communication device  604  receives the entire packet  504 , but only processes those frequency bins  502  and/or timeslots  506  that are assigned to it. Preferably, each device  604  is assigned non-adjacent frequency bins  502 , as in  FIG. 5 . This way, if interference corrupts the information in a portion of communication channel  500 , then the effects are spread across all devices  604  within a cell  602 . Hopefully, by spreading out the effects of interference in this manner the effects are minimized and the entire information sent to each device  604  can still be recreated from the unaffected information received in other frequency bins. For example, if interference, such as fading, corrupted the information in bins f 0 -f 4 , then each user  1 - 4  loses one packet of data. But each user potentially receives three unaffected packets from the other bins assigned to them. Hopefully, the unaffected data in the other three bins provides enough information to recreate the entire message for each user. Thus, frequency diversity can be achieved by assigning non-adjacent bins to each of multiple users.  
         [0087]     Ensuring that the bins assigned to one user are separated by more than the coherence bandwidth ensures frequency diversity. As discussed above, the coherence bandwidth is approximately equal to 1/d s . For outdoor systems, where ds is typically 1 microsecond, 1/d s ={fraction (1/1)} microsecond=1 megahertz (MHz). Thus, the non-adjacent frequency bands assigned to a user are preferably separated by at least 1 MHz. It can be even more preferable, however, if the coherence bandwidth plus some guard band to ensure sufficient frequency diversity separate the non-adjacent bins assigned to each user. For example, it is preferable in certain implementations to ensure that at least 5 times the coherence bandwidth, or 5 MHz in the above example, separates the non-adjacent bins.  
         [0088]     Another way to provide frequency diversity is to repeat blocks of data in frequency bins assigned to a particular user that are separated by more than the coherence bandwidth. In other words, if 4 sub-channels  200  are assigned to a user, then data block a can be repeated in the first and third sub-channels  200  and data block b can be repeated in the second and fourth sub-channels  202 , provided the sub-channels are sufficiently separated in frequency. In this case, the system can be said to be using a diversity length factor of 2. The system can similarly be configured to implement other diversity lengths, e.g., 3, 4, . . . , l.  
         [0089]     It should be noted that spatial diversity can also be included depending on the embodiment. Spatial diversity can comprise transmit spatial diversity, receive spatial diversity, or both. In transmit spatial diversity, the transmitter uses, a plurality of separate transmitters and a plurality of separate antennas to transmit each message. In other words, each transmitter transmits the same message in parallel. The messages are then received from the transmitters and combined in the receiver. Because the parallel transmissions travel different paths, if one is affected by fading, the others will likely not be affected. Thus, when they are combined in the receiver, the message should be recoverable even if one or more of the other transmission paths experienced severe fading.  
         [0090]     Receive spatial diversity uses a plurality of separate receivers and a plurality of separate antennas to receive a single message. If an adequate distance separates the antennas, then the transmission path for the signals received by the antennas will be different. Again, this difference in the transmission paths will provide imperviousness to fading when the signals from the receivers are combined.  
         [0091]     Transmit and receive spatial diversity can also be combined within a system such as system  600  so that two antennas are used to transmit and two antennas are used to receive. Thus, each base station  606  transmitter can include two antennas, for transmit spatial diversity, and each communication device  604  receiver can include two antennas, for receive spatial diversity. If only transmit spatial diversity is implemented in system  600 , then it can be implemented in base stations  606  or in communication devices  604 . Similarly, if only receive spatial diversity is included in system  600 , then it can be implemented in base stations  606  or communication devices  604 .  
         [0092]     The number of communication devices  604  assigned frequency bins  502  and/or time slots  506  in each cell  602  is preferably programmable in real time. In other words, the resource allocation within a communication cell  602  is preferably programmable in the face of varying external conditions, i.e., multipath or adjacent cell interference, and varying requirements, i.e., bandwidth requirements for various users within the cell. Thus, if user  1  requires the whole bandwidth to download a large video file, for example, then the allocation of bins  502  can be adjusted to provide user  1  with more, or even all, of bins  502 . Once user  1  no longer requires such large amounts of bandwidth, the allocation of bins  502  can be readjusted among all of users  1 - 4 .  
         [0093]     It should also be noted that all of the bins assigned to a particular user can be used for both the forward and reverse link. Alternatively, some bins  502  can be assigned as the forward link and some can be assigned for use on the reverse link, depending on the implementation.  
         [0094]     To increase capacity, the entire bandwidth B is preferably reused in each communication cell  602 , with each cell  602  being differentiated by a unique synchronization code (see discussion below). Thus, system  600  provides increased immunity to multipath and fading as well as increased band width due to the elimination of frequency reuse requirements.  
         [heading-0095]     3. Synchronization  
         [0096]      FIG. 8  illustrates an example embodiment of a synchronization code correlator  800 . When a device  604  in cell  1  (see  FIG. 6 ), for example, receives an incoming communication from the cell  1  base station  606 , it compares the incoming data with SYNC 1  in correlator  800 . Essentially, the device scans the incoming data trying to correlate the data with the known synchronization code, in this case SYNC 1 . Once correlator  800  matches the incoming data to SYNC 1  it generates a correlation peak  804  at the output. Multipath versions of the data will also generate correlation peaks  806 , although these peaks  806  are generally smaller than correlation peak  804 , The device can then use the correlation peaks to perform channel estimation, which allows the device to adjust for the multipath using, e.g., an equalizer. Thus, in cell  1 , if correlator  800  receives a data stream comprising SYNC 1 , it will generate correlation peaks  804  and  806 . If, on the other hand, the data stream comprises SYNC 2 , for example, then no peaks will be generated and the device will essentially ignore the incoming communication.  
         [0097]     Even though a data stream that comprises SYNC 2  will not create any correlation peaks, it can create noise in correlator  800  that can prevent detection of correlation peaks  804  and  806 . Several steps can be taken to prevent this from occurring. One way to minimize the noise created in correlator  800  by signals from adjacent cells  602 , is to configure system  600  so that each base station  606  transmits at the same time. This way, the synchronization codes can preferably be generated in such a manner that only the synchronization codes  704  of adjacent cell data streams, e.g., streams  708 ,  710 , and  712 , as opposed to packets  702  within those streams, will interfere with detection of the correct synchronization code  704 , e.g., SYNC 1 . The synchronization codes can then be further configured to eliminate or reduce the interference.  
         [0098]     For example, the noise or interference caused by an incorrect synchronization code is a function of the cross correlation of that synchronization code with respect to the correct code. The better the cross correlation between the two, the lower the noise level. When the cross correlation is ideal, then the noise level will be virtually zero as illustrated in  FIG. 9  by noise level  902 . Therefore, a preferred embodiment of system  600  uses synchronization codes that exhibit ideal cross correlation, i.e., zero. Preferably, the ideal cross correlation of the synchronization codes covers a period  1  that is sufficient to allow accurate detection of multipath correlation peaks  906  as well as correlation peak  904 . This is important so that accurate channel estimation and equalization can take place. Outside of period  1 , the noise level  908  goes up, because the data in packets  702  is random and will exhibit low cross correlation with the synchronization code, e.g., SYNC 1 . Preferably, period  1  is actually slightly longer then the multipath length in order to ensure that the multipath can be detected.  
         [0099]     a. Synchronization Code Generation  
         [0100]     Conventional systems use orthogonal codes to achieve cross correlation in correlator  800 . In system  600  for example, SYNC 1 , SYNC 2 , SYNC 3 , and SYNC 4 , corresponding to cells  1 - 4  (see lightly shaded cells  602  of  FIG. 6 ) respectively, will all need to be generated in such a manner that they will have ideal cross correlation with each other. In one embodiment, if the data streams involved comprise high and low data bits, then the value “1” can be assigned to the high data bits and “−1” to the low data bits. Orthogonal data sequences are then those that produce a “0” output when they are exclusively ORed (XORed) together in correlator  800 . The following example illustrates this point for orthogonal sequences  1  and  2 :  
               sequence   ⁢           ⁢   1   ⁢     :             11   -   11               sequence   ⁢           ⁢   2   ⁢     :             111   -   1           _       
               ⁢         1   ⁢           ⁢   1     -   1   -   1     =   0         
 
         [0101]     Thus, when the results of XORing each bit pair are added, the result is “0”.  
         [0102]     But in system  600 , for example, each code must have ideal, or zero, cross correlation with each of the other codes used in adjacent cells  602 . Therefore, in one example embodiment of a method for generating synchronization codes exhibiting the properties described above, the process begins by selecting a “perfect sequence” to be used as the basis for the codes. A perfect sequence is one that when correlated with itself produces a number equal to the number of bits in the sequence. For example:  
         Perfect   ⁢           ⁢   sequence   ⁢           ⁢   1   ⁢     :     ⁢           ⁢   11     -   11       
               ⁢       11   -   11     _         
               ⁢       1   ⁢           ⁢   1   ⁢           ⁢   1   ⁢           ⁢   1     =   4         
 
         [0103]     But each time a perfect sequence is cyclically shifted by one bit, the new sequence is orthogonal with the original sequence. Thus, for example, if perfect sequence  1  is cyclically shifted by one bit and then correlated with the original, the correlation produces a “0” as in the following example:  
         Perfect   ⁢           ⁢   sequence   ⁢           ⁢   1   ⁢     :     ⁢           ⁢   11     -   11       
               ⁢         1   ⁢           ⁢   1   ⁢           ⁢   1     -   1     _         
               ⁢         1   ⁢           ⁢   1     ⁢           -   1   -   1     =   0         
 
         [0104]     If the perfect sequence  1  is again cyclically shifted by one bit, and again correlated with the original, then it will produce a “0”. In general, you can cyclically shift a perfect sequence by any number of bits up to its length and correlate the shifted sequence with the original to obtain a “0”.  
         [0105]     Once a perfect sequence of the correct length is selected, the first synchronization code is preferably generated in one embodiment by repeating the sequence 4 times. Thus, if perfect sequence  1  is being used, then a first synchronization code y would be the following: 
 
 y =1 1 −1 1 1 1 −1 1 1 1 −1 1 1 1 −1 1. 
 
         [0106]     Or in generic form: 
 
 y=x (0) x (1) x (2) x (3) x (0) x (1) x (2) x (3) x (0) x (1) x (2) x (3) x (0) x (1) x (2) x (3). 
 
         [0107]     For a sequence of length L: 
 
 y=x (0) x (1) . . .  x ( L ) x (0) x (1) . . .  x ( L ) x (0) x (1) . . .  x ( L ) x (0) x (1) . . . 
 
         [0108]     Repeating the perfect sequence allows correlator  800  a better opportunity to detect the synchronization code and allows generation of other uncorrelated frequencies as well. Repeating has the effect of sampling in the frequency domain. This effect is illustrated by the graphs in  FIG. 10 . Thus, in trace  1 , which corresponds to synchronization code y, a sample  1002  is generated every fourth sample bin  1000 . Each sample bin is separated by 1/(4L×T), where T is the symbol duration. Thus in the above example, where L=4, each sample bin is separated by l/(16×7) in the frequency domain. Traces  2 - 4  illustrate the next three synchronization codes. As can be seen, the samples for each subsequent synchronization code are shifted by one sample bin relative to the samples for the previous sequence. Therefore, none of sequences interfere with each other.  
         [0109]     To generate the subsequent sequences, corresponding to traces  2 - 4 , sequence y must be shifted in frequency. This can be accomplished using the following equation: 
 
 z   r ( m )= y ( m )*exp( j* 2 *π*r*m /( n*L )),  (5) 
        for r=1 to L (# of sequences) and m=0 to 4*L−1 (time); and     where: z r (m)=each subsequent sequence; 
            y(m)=the first sequence; and     n=the number of times the sequence is repeated.    
               
 
         [0114]     It will be understood that multiplying by an exp(j2π(r*m/N)) factor, where N is equal to the number of times the sequence is repeated (n) multiplied by the length of the underlying perfect sequence Z, in the time domain results in a shift in the frequency domain. Equation (5) results in the desired shift as illustrated in  FIG. 10  for each of synchronization codes  2 - 4 , relative to synchronization code  1 . The final step in generating each synchronization code is to append the copies of the last M samples, where M is the length of the multipath, to the front of each code. This is done to make the convolution with the multipath cyclic and to allow easier detection of the multipath.  
         [0115]     It should be noted that synchronization codes can be generated from more than one perfect sequence using the same methodology. For example, a perfect sequence can be generated and repeated for times and then a second perfect sequence can be generated and repeated four times to get a n factor equal to eight. The resulting sequence can then be shifted as described above to create the synchronization codes.  
         [0116]     b. Signal Measurements Using Synchronization Codes  
         [0117]     Therefore, when a communication device is at the edge of a cell, it will receive signals from multiple base stations and, therefore, will be decoding several synchronization codes at the same time. This can be illustrated with the help of  FIG. 11 , which illustrates another example embodiment of a wireless communication system  1100  comprising communication cells  1102 ,  1104 , and  1106  as well as communication device  1108 , which is in communication with base station  1110  of cell  1102  but also receiving communication from base stations  1112  and  1114  of cells  1104  and  1106 , respectively.  
         [0118]     If communications from base station  1110  comprise synchronization code SYNC 1  and communications from base station  1112  and  1114  comprise SYNC 2  and SYNC 3  respectively, then device  1108  will effectively receive the sum of these three synchronization codes. This is because, as explained above, base stations  1110 ,  1112 , and  1114  are configured to transmit at the same time. Also, the synchronization codes arrive at device  1108  at almost the same time because they are generated in accordance with the description above.  
         [0119]     Again as described above, the synchronization codes SYNC 1 , SYNC 2 , and SYNC 3  exhibit ideal cross correlation. Therefore, when device  1108  correlates the sum x of codes SYNC 1 , SYNC 2 , and SYNC 3 , the latter two will not interfere with proper detection of SYNC 1  by device  1108 . Importantly, the sum x can also be used to determine important signal characteristics, because the sum x is equal to the sum of the synchronization code signal in accordance with the following equation: 
 
 x =SYNC 1 +SYNC 2 +SYNC 3 .  (6) 
 
         [0120]     Therefore, when SYNC 1  is removed, the sum of SYNC 2  and SYNC 3  is left, as shown in the following: 
 
 x −SYNC 1 =SYNC 2 +SYNC 3 .  (7) 
 
 The energy computed from the sum (SYNC 2 +SYNC 3 ) is equal to the noise or interference seen by device  1108 . Since the purpose of correlating the synchronization code in device  1106  is to extract the energy in SYNC 1 , device  1108  also has the energy in the signal from base station  1110 , i.e., the energy represented by SYNC 1 . Therefore, device  1106  can use the energy of SYNC 1  and of (SYNC 2 +SYNC 3 ) to perform a signal-to-interference measurement for the communication channel over which it is communicating with base station  1110 . The result of the measurement is preferably a signal-to-interference ratio (SIR). The SIR measurement can then be communicated back to base station  1110  for purposes that will be discussed below. 
 
         [0122]     The ideal cross correlation of the synchronization codes also allows device  1108  to perform extremely accurate determinations of the Channel Impulse Response (CIR), or channel estimation, from the correlation produced by correlator  800 . This allows for highly accurate equalization using low cost, low complexity equalizers, thus overcoming a significant draw back of conventional systems.  
         [heading-0123]     4. Sub-Channel Assignments  
         [0124]     As mentioned, the SIR as determined by device  1108  can be communicated back to base station  1110  for use in the assignment of slots  502 . In one embodiment, due to the fact that each sub-channel  502  is processed independently, the SIR for each sub-channel  502  can be measured and communicated back to base station  1110 . In such an embodiment, therefore, sub-channels  502  can be divided into groups and a SIR measurement for each group can be sent to base station  1110 . This is illustrated in  FIG. 12A , which shows a wideband communication channel  1200  segmented into sub-channels f 0  to f 15 . Sub-channels f 0  to f 15  are then grouped into 8 groups G 1  to G 8 . Thus, in one embodiment, device  1108  and base station  1110  communicate over a channel such as channel  1200 .  
         [0125]     Sub-channels in the same group are preferably separated by as many sub-channels as possible to ensure diversity. In  FIG. 12A  for example, sub-channels within the same group are 7 sub-channels apart, e.g., group G 1  comprises f 0  and f 8 .  
         [0126]     Device  1102  reports a SIR measurement for each of the groups G 1  to G 8 . These SIR measurements are preferably compared with a threshold value to determine which sub-channels groups are useable by device  1108 . This comparison can occur in device  1108  or base station  1110 . If it occurs in device  1108 , then device  1108  can simply report to base station  1110  which sub-channel groups are useable by device  1108 .  
         [0127]     SIR reporting will be simultaneously occurring for a plurality of devices within cell  1102 . Thus,  FIG. 12B  illustrates the situation where two communication devices corresponding to user 1  and user 2  report SIR levels above the threshold for groups G 1 , G 3 , G 5 , and G 7 . Base station  1110  preferably then assigns sub-channel groups to user 1  and user 2  based on the SIR reporting as illustrated in  FIG. 12B . When assigning the “good” sub-channel groups to user 1  and user 2 , base station  1110  also preferably assigns them based on the principles of frequency diversity. In  FIG. 12B , therefore, user 1  and user 2  are alternately assigned every other “good” sub-channel.  
         [0128]     The assignment of sub-channels in the frequency domain is equivalent to the assignment of time slots in the time domain. Therefore, as illustrated in  FIG. 13 , two users, user 1  and user 2 , receive packet  1302  transmitted over communication channel  1200 .  FIG. 13  also illustrated the sub-channel assignment of  FIG. 12B . While  FIGS. 12 and 13  illustrate sub-channel/time slot assignment based on SIR for two users, the principles illustrated can be extended for any number of users. Thus, a packet within cell  1102  can be received by 3 or more users. Although, as the number of available sub-channels is reduced due to high SIR, so is the available bandwidth. In other words, as available sub-channels are reduced, the number of users that can gain access to communication channel  1200  is also reduced.  
         [0129]     Poor SIR can be caused for a variety of reasons, but frequently it results from a device at the edge of a cell receiving communication signals from adjacent cells. Because each cell is using the same bandwidth B, the adjacent cell signals will eventually raise the noise level and degrade SIR for certain sub-channels. In certain embodiments, therefore, sub-channel assignment can be coordinated between cells, such as cells  1102 ,  1104 , and  1106  in  FIG. 11 , in order to prevent interference from adjacent cells.  
         [0130]     Thus, if communication device  1108  is near the edge of cell  1102 , and device  1118  is near the edge of cell  1106 , then the two can interfere with each other. As a result, the SIR measurements that device  1108  and  1118  report back to base stations  1110  and  1114 , respectively, will indicate that the interference level is too high. Base station  1110  can then be configured to assign only the odd groups, i.e., G 1 , G 3 , G 5 , etc., to device  1108 , while base station  1114  can be configured to assign the even groups to device  1118  in a coordinated fashion. The two devices  1108  and  1118  will then not interfere with each other due to the coordinated assignment of sub-channel groups.  
         [0131]     Assigning the sub-channels in this manner reduces the overall bandwidth available to devices  1108  and  1118 , respectively. In this case the bandwidth is reduced by a factor of two. But it should be remembered that devices operating closer to each base station  1110  and  1114 , respectively, will still be able to use all sub-channels if needed. Thus, it is only devices, such as device  1108 , that are near the edge of a cell that will have the available bandwidth reduced. Contrast this with a CDMA system, for example, in which the bandwidth for all users is reduced, due to the spreading techniques used in such systems, by approximately a factor of 10 at all times. It can be seen, therefore, that the systems and methods for wireless communication over a wide bandwidth channel using a plurality of sub-channels not only improves the quality of service, but can also increase the available bandwidth significantly.  
         [0132]     When there are three devices  1108 ,  1118 , and  1116  near the edge of their respective adjacent cells  1102 ,  1104 , and  1106 , the sub-channels can be divided by three. Thus device  1108 , for example, can be assigned groups G 1 , G 4 , etc., device  1118  can be assigned groups G 2 , G 5 , etc., and device  1116  can be assigned groups G 3 , G 6 , etc. In this case the available bandwidth for these devices, i.e., devices near the edges of cells  1102 ,  1104 , and  1106 , is reduced by a factor of 3, but this is still better than a CDMA system, for example.  
         [0133]     The manner in which such a coordinated assignment of sub-channels can work is illustrated by the flow chart in  FIG. 14 . First in step  1402 , a communication device, such as device  1108 , reports the SIR for all sub-channel groups G 1  to G 8 . The SIRs reported are then compared, in step  1404 , to a threshold to determine if the SIR is sufficiently low for each group. Alternatively, device  1108  can make the determination and simply report which groups are above or below the SIR threshold. If the SIR levels are good for each group, then base station  1110  can make each group available to device  1108 , in step  1406 . Periodically, device  1108  preferably measures the SIR level and updates base station  1110  in case the SIR as deteriorated. For example, device  1108  may move from near the center of cell  1102  toward the edge, where interference from an adjacent cell may affect the SIR for device  1108 .  
         [0134]     If the comparison in step  1404  reveals that the SIR levels are not good, then base station  1110  can be preprogrammed to assign either the odd groups or the even groups only to device  1108 , which it will do in step  1408 . Device  1108  then reports the SIR measurements for the odd or even groups it is assigned in step  1410 , and they are again compared to a SIR threshold in step  1412 .  
         [0135]     It is assumed that the poor SIR level is due to the fact that device  1108  is operating at the edge of cell  1102  and is therefore being interfered with by a device such as device  1118 . But device  1108  will be interfering with device  1118  at the same time. Therefore, the assignment of odd or even groups in step  1408  preferably corresponds with the assignment of the opposite groups to device  1118 , by base station  1114 , Accordingly, when device  1108  reports the SIR measurements for whichever groups, odd or even, are assigned to it, the comparison in step  1410  should reveal that the SIR levels are now below the threshold level. Thus, base station  1110  makes the assigned groups available to device  1108  in step  1414 . Again, device  1108  preferably periodically updates the SIR measurements by returning to step  1402 .  
         [0136]     It is possible for the comparison of step  1410  to reveal that the SIR levels are still above the threshold, which should indicate that a third device, e.g., device  1116  is still interfering with device  1108 . In this case, base station  1110  can be preprogrammed to assign every third group to device  1108  in step  1416 . This should correspond with the corresponding assignments of non-interfering channels to devices  1118  and  1116  by base stations  1114  and  1112 , respectively. Thus, device  1108  should be able to operate on the sub-channel groups assigned, i.e., G 1 , G 4 , etc., without undue interference. Again, device  1108  preferably periodically updates the SIR measurements by returning to step  1402 . Optionally, a third comparison step (not shown) can be implemented after step  1416 , to ensure that the groups assigned to device  1408  posses an adequate SIR level for proper operation. Moreover, if there are more adjacent cells, i.e., if it is possible for devices in a 4 th  or even a 5 th  adjacent cell to interfere with device  1108 , then the process of  FIG. 14  would continue and the sub-channel groups would be divided even further to ensure adequate SIR levels on the sub-channels assigned to device  1108 .  
         [0137]     Even though the process of  FIG. 14  reduces the bandwidth available to devices at the edge of cells  1102 ,  1104 , and  1106 , the SIR measurements can be used in such a manner as to increase the data rate and therefore restore or even increase bandwidth. To accomplish this, the transmitters and receivers used in base stations  1102 ,  1104 , and  1106 , and in devices in communication therewith, e.g., devices  1108 ,  1114 , and  1116  respectively, must be capable of dynamically changing the symbol mapping schemes used for some or all of the sub-channel. For example, in some embodiments, the symbol mapping scheme can be dynamically changed among BPSK, QPSK, 8PSK, 16QAM, 32QAM, etc. As the symbol mapping scheme moves higher, i.e., toward 32QAM, the SIR level required for proper operation moves higher, i.e., less and less interference can be withstood. Therefore, once the SIR levels are determined for each group, the base station, e.g., base station  1110 , can then determine what symbol mapping scheme can be supported for each sub-channel group and can change the modulation scheme accordingly. Device  1108  must also change the symbol mapping scheme to correspond to that of the base stations. The change can be effected for all groups uniformly, or it can be effected for individual groups. Moreover, the symbol mapping scheme can be changed on just the forward link, just the reverse link, or both, depending on the embodiment.  
         [0138]     Thus, by maintaining the capability to dynamically change the symbol mapping scheme used for assigned sub-channels, the systems and methods described herein provide the ability to maintain higher available bandwidths with higher performance levels than conventional systems. To fully realize the benefits described, however, the systems and methods described thus far must be capable of implementation in a cost effect and convenient manner. Moreover, the implementation must include reconfigurability so that a single device can move between different types of communication systems and still maintain optimum performance in accordance with the systems and methods described herein. The following descriptions detail example high level embodiments of hardware implementations configured to operate in accordance with the systems and methods described herein in such a manner as to provide the capability just described above.  
         [heading-0139]     5. Sample Transmitter Embodiments  
         [0140]      FIG. 15  is logical block diagram illustrating an example embodiment of a transmitter  1500  configured for wireless communication in accordance with the systems and methods described above. The transmitter could, for example be within a base station, e.g., base station  606 , or within a communication device, such as device  604 . Transmitter  1500  is provided to illustrate logical components that can be included in a transmitter configured in accordance with the systems and methods described herein. It is not intended to limit the systems and methods for wireless communication over a wide bandwidth channel using a plurality of sub-channels to any particular transmitter configuration or any particular wireless communication system.  
         [0141]     With this in mind, it can be seen that transmitter  1500  comprises a serial-to-parallel converter  1504  configured to receive a serial data stream  1502  comprising a data rate R. Serial-to-parallel converter  1504  converts data stream  1502  into N parallel data streams  1504 , where N is the number of sub-channels  200 . It should be noted that while the discussion that follows assumes that a single serial data stream is used, more than one serial data stream can also be used if required or desired. In any case, the data rate of each parallel data stream  1504  is then R/N. Each data stream  1504  is then sent to a scrambler, encoder, and interleaver block  1506 . Scrambling, encoding, and interleaving are common techniques implemented in many wireless communication transmitters and help to provide robust, secure communication. Examples of these techniques will be briefly explained for illustrative purposes.  
         [0142]     Scrambling breaks up the data to be transmitted in an effort to smooth out the spectral density of the transmitted data. For example, if the data comprises a long string of “1”s, there will be a spike in the spectral density. This spike can cause greater interference within the wireless communication system. By breaking up the data, the spectral density can be smoothed out to avoid any such peaks. Often, scrambling is achieved by XORing the data with a random sequence.  
         [0143]     Encoding, or coding, the parallel bit streams  1504  can, for example, provide Forward Error Correction (FEC). The purpose of FEC is to improve the capacity of a communication channel by adding some carefully designed redundant information to the data being transmitted through the channel. The process of adding this redundant information is known as channel coding. Convolutional coding and block coding are the two major forms of channel coding. Convolutional codes operate on serial data, one or a few bits at a time. Block codes operate on relatively large (typically, up to a couple of hundred bytes) message blocks. There are a variety of useful convolutional and block codes, and a variety of algorithms for decoding the received coded information sequences to recover the original data. For example, convolutional encoding or turbo coding with Viterbi decoding is a FEC technique that is particularly suited to a channel in which the transmitted signal is corrupted mainly by additive white gaussian noise (AWON) or even a channel that simply experiences fading.  
         [0144]     Convolutional codes are usually described using two parameters: the code rate and the constraint length. The code rate, k/n, is expressed as a ratio of the number of bits into the convolutional encoder (k) to the number of channel symbols (n) output by the convolutional encoder in a given encoder cycle. A common code rate is ½, which means that 2 symbols are produced for every 1-bit input into the coder. The constraint length parameter, K, denotes the “length” of the convolutional encoder, i.e. how many &amp;-bit stages are available to feed the combinatorial logic that produces the output symbols. Closely related to K is the parameter m, which indicates how many encoder cycles an input bit is retained and used for encoding after it first appears at the input to the convolutional encoder. The m parameter can be thought of as the memory length of the encoder.  
         [0145]     Interleaving is used to reduce the effects of fading. Interleaving mixes up the order of the data so that if a fade interferes with a portion of the transmitted signal, the overall message will not be effected. This is because once the message is de-interleaved and decoded in the receiver, the data lost wall comprise non-contiguous portions of the overall message. In other words, the fade will interfere with a contiguous portion of the interleaved message, but when the message is de-interleaved, the interfered with portion is spread throughout the overall message. Using techniques such as FEC, the missing information can then be filled in, or the impact of the lost data may just be negligible.  
         [0146]     After blocks  1506 , each parallel data stream  1504  is sent to symbol mappers  1508 . Symbol mappers  1508  apply the requisite symbol mapping, e.g., BPSK, QPSK. etc.; to each parallel data stream  1504 . Symbol mappers  1508  are preferably programmable so that the modulation applied to parallel data streams can be changed, for example, in response to the SIR reported for each sub-channel  202 . It is also preferable, that each symbol mapper  1508  be separately programmable so that the optimum symbol mapping scheme for each sub-channel can be selected and applied to each parallel data stream  1504 .  
         [0147]     After symbol mappers  1508 , parallel data streams  1504  are sent to modulators  1510 . Important aspects and features of example embodiments of modulators  1510  are described below. After modulators  1510 , parallel data streams  1504  are sent to summer  1512 , which is configured to sum the parallel data streams and thereby generate a single serial data stream  1518  comprising each of the individually processed parallel data streams  1504 . Serial data stream  1518  is then sent to radio module  1512 , where it is modulated with an RJF carrier, amplified, and transmitted via antenna  1516  according to known techniques. Radio module embodiments that can be used in conjunction with the systems and methods described herein are described below.  
         [0148]     The transmitted signal occupies the entire bandwidth B of communication channel  100  and comprises each of the discrete parallel data streams  1504  encoded onto their respective sub-channels  102  within bandwidth B. Encoding parallel data streams  1504  onto the appropriate sub-channels  102  requires that each parallel data stream  1504  be shifted in frequency by an appropriate offset. This is achieved in modulator  1510 .  
         [0149]      FIG. 16  is a logical block diagram of an example embodiment of a modulator  1600  in accordance with the systems and methods described herein. Importantly, modulator  1600  takes parallel data streams  1602  performs Time Division Modulation (TDM) or Frequency Division Modulation (FDM) on each data stream  1602 , filters them using filters  1612 , and then shifts each data stream in frequency using frequency shifter  1614  so that they occupy the appropriate sub-channel. Filters  1612  apply the required pulse snapping, i.e., they apply the roll-off factor described in section  1 . The frequency shifted parallel data streams  1602  are then summed and transmitted. Modulator  1600  can also include rate controller  1604 , frequency encoder  1606 , and interpolators  1610 . All of the components shown in  FIG. 16  are described in more detail in the following paragraphs and in conjunction with  FIGS. 17-23 .  
         [0150]      FIG. 17  illustrates one example embodiment of a rate controller  1700  in accordance with the systems and methods described herein. Rate control  1700  is used to control the data rate of each parallel data stream  1602 . In rate controller  1700 , the data rate is halved by repeating data streams d( 0 ) to d( 7 ), for example, producing streams a( 0 ) to a( 15 ) in which a( 0 ) is the same as a( 8 ), a( 1 ) is the same as a( 9 ), etc.  FIG. 17  illustrates that the effect of repeating the data streams in this manner is to take the data streams that are encoded onto the first  8  sub-channels  1702 , and duplicate them on the next 8 sub-channels  1702 . As can be seen, 7 sub-channels separate sub-channels  1702  comprising the same, or duplicate, data streams. Thus, if fading effects one sub-channel  1702 , for example, the other sub-channels  1702  carrying the same data will likely not be effected, i.e., there is frequency diversity between the duplicate data streams. So by sacrificing data rate, in this case half the data rate, more robust transmission is achieved. Moreover, the robustness provided by duplicating the data streams d( 0 ) to d( 8 ) can be further enhanced by applying scrambling to the duplicated data streams via scramblers  1704 .  
         [0151]     It should be noted that the data rate can be reduced by more than half, e.g. by four or more. Alternatively, the data rate can also be reduced by an amount other than half. For example if information from n data stream is encoded m sub-channels, where m&gt;n. Thus, to decrease the rate by ⅔, information from one data stream can be encoded on a first sub-channel, information from a second data stream can be encoded on a second data channel, and the sum or difference of the two data streams can be encoded on a third channel. In which case, proper scaling will need to be applied to the power in the third channel. Otherwise, for example, the power in the third channel can be twice the power in the first two.  
         [0152]     Preferably, rate controller  1700  is programmable so that the data rate can be changed responsive to certain operational factors. For example, if the SIR reported for sub-channels  1702  is low, then rate controller  1700  can be programmed to provide more robust transmission via repetition to ensure that no data is lost due to interference. Additionally, different types of wireless communication system, e.g., indoor, outdoor, line-of-sight, may require varying degrees of robustness. Thus, rate controller  1700  can be adjusted to provide the minimum required robustness for the particular type of communication system. This type of programmability not only ensures robust communication, it can also be used to allow a single device to move between communication systems and maintain superior performance.  
         [0153]      FIG. 18  illustrates an alternative example embodiment of a rate controller  1800  in accordance with the systems and methods described. In rate controller  1800  the data rate is increased instead of decreased. This is accomplished using serial-to-parallel converters  1802  to convert each data streams d( 0 ) to d( 15 ), for example, into two data streams. Delay circuits  1804  then delay one of the two data streams generated by each serial-to-parallel converter  1802  by ½ a symbol. Thus, data streams d( 0 ) to d( 15 ) are transformed into data streams a( 0 ) to a( 31 ). The data streams generated by a particular serial-to-parallel converter  1802  and associate delay circuit  1804  must then be summed and encoded onto the appropriate sub-channel. For example, data streams a( 0 ) and a(l) must be summed and encoded onto the first sub-channel. Preferably, the data streams are summed subsequent to each data stream being pulsed shaped by a filter  1612 .  
         [0154]     Thus, rate controller  1604  is preferably programmable so that the data rate can be increased, as in rate controller  1800 , or decreased, as in rate controller  1700 , as required by a particular type of wireless communication system, or as required by the communication channel conditions or sub-channel conditions. In the event that the data rate is increased, filters  1612  are also preferably programmable so that they can be configured to apply pulse snapping to data streams a( 0 ) to a( 31 ), for example, and then sum the appropriate streams to generate the appropriate number of parallel data streams to send to frequency shifter  1614 .  
         [0155]     The advantage of increasing the data rate in the manner illustrated in  FIG. 18  is that higher symbol mapping rates can essentially be achieved, without changing the symbol mapping used in symbol mappers  1508 . Once the data streams are summed, the summed streams are shifted in frequency so that they reside in the appropriate sub-channel But because the number of bits per each symbol has been doubled, the symbol mapping rate has been doubled. Thus, for example, a 4QAM symbol mapping can be converted to a 16QAM symbol mapping, even if the SIR is too high for 16QAM symbol mapping to otherwise be applied. In other words, programming rate controller  1800  to increase the data rate in the manner illustrated in  FIG. 18  can increase the symbol mapping even when channel conditions would otherwise not allow it, which in turn can allow a communication device to maintain adequate or even superior performance regardless of the type of communication system.  
         [0156]     The draw back to increasing the data rate as illustrated in  FIG. 18  is that interference is increased, as is receiver complexity. The former is due to the increased amount of data. The latter is due to the fact that each symbol cannot be processed independently because of the ½ symbol overlap. Thus, these concerns must be balanced against the increase symbol mapping ability when implementing a rate controller such as rate controller  1800 .  
         [0157]      FIG. 19  illustrates one example embodiment of a frequency encoder  1900  in accordance with the systems and methods described herein. Similar to rate encoding, frequency encoding is preferably used to provide increased communication robustness. In frequency encoder  1900  the sum or difference of multiple data streams are encoded onto each sub-channel. This is accomplished using adders  1902  to sum data streams d( 0 ) to d( 7 ) with data streams d( 8 ) to d( 15 ), respectively, while adders  1904  subtract data streams d( 0 ) to d( 7 ) from data streams d( 8 ) to d( 15 ), respectively, as shown. Thus, data streams a( 0 ) to a( 15 ) generated by adders  1902  and  1904  comprise information related to more than one data streams d( 0 ) to d( 15 ). For example, a( 0  comprises the sum of d( 0 ) and d( 8 ), i.e., d( 0 )+d( 8 ), while a( 8 ) comprises d( 8 ) d( 0 ). Therefore, if either a( 0 ) or a( 8 ) is not received due to fading, for example, then both of data streams d( 0 ) and d( 8 ) can still be retrieved from data stream a( 8 ).  
         [0158]     Essentially, the relationship between data stream d( 0 ) to d( 15 ) and a( 0 ) to a( 15 ) is a matrix relationship. Thus, if the receiver knows the correct matrix to apply, it can recover the sums and differences of d( 0 ) to d( 15 ) from a( 0 ) to a( 15 ). Preferably, frequency encoder  1900  is programmable, so that it can be enabled and disabled in order to provided robustness when required. Preferable, adders  1902  and  1904  are programmable also so that different matrices can be applied to d( 0 ) to d( 15 ).  
         [0159]     After frequency encoding, if it is included, data streams  1602  are sent to TDM/TDM blocks  1608 . TDM/FDM blocks  1608  perform TDM or FDM on the data streams as required by the particular embodiment.  FIG. 20  illustrates an example embodiment of a TDM/FDM block  2000  configured to perform TDM on a data stream. TDM/FDM block  2000  is provided to illustrate the logical components that can be included in a TDM/FDM block configured to perform TDM on a data stream, Depending on the actual implementation, some of the logical components may or may not be included. TDM/FDM block  2000  comprises a sub-block repeater  2002 , a sub-block scrambler  2004 , a sub-block terminator  2006 , a sub-block repeater  2008 , and a SYNC inserter  2010 .  
         [0160]     Sub-block repeater  2002  is configured to receive a sub-block of data, such as block  2012  comprising bits a( 0 ) to a( 3 ) for example. Sub-block repeater is then configured to repeat block  2012  to provide repetition, which in turn leads to more robust communication. Thus, sub-block repeater  2002  generates block  2014 , which comprises 2 blocks  2012 . Sub-block scrambler  2004  is then configured to receive block  2014  and to scramble it, thus generating block  2016 . One method of scrambling can be to invert half of block  2014  as illustrated in block  2016 . But other scrambling methods can also be implemented depending on the embodiment.  
         [0161]     Sub-block terminator  2006  takes block  2016  generated by sub-block scrambler  2004  and adds a termination block  2034  to the front of block  2016  to form block  2018 . Termination block  2034  ensures that each block can be processed independently in the receiver. Without termination block  2034 , some blocks may be delayed due to multipath, for example, and they would therefore overlap part of the next block of data. But by including termination block  2034 , the delayed block can be prevented from overlapping any of the actual data in the next block.  
         [0162]     Termination block  2034  can be a cyclic prefix termination  2036 . A cyclic prefix termination  2036  simply repeats the last few symbols of block  2018 . Thus, for example, if cyclic prefix termination  2036  is three symbols long, then it would simply repeat the last three symbols of block  2018 . Alternatively, termination block  2034  can comprise a sequence of symbols that are known to both the transmitter and receiver. The selection of what type of block termination  2034  to use can impact what type of equalizer is used in the receiver. Therefore, receiver complexity and choice of equalizers must be considered when determining what type of termination block  2034  to use in TDM/FDM block  2000 .  
         [0163]     After sub-block terminator  2006 , TDM/FDM block  2000  can include a sub-block repeater  2008  configured to perform a second block repetition step in which block  2018  is repeated to form block  2020 . in certain embodiments, sub-block repeater can be configured to perform a second block scrambling step as well. After sub-block repeater  2008 , if included, TDM/FDM block  2000  comprises a SYNC inserter  210  configured to periodically insert an appropriate synchronization code  2032  after a predetermined number of blocks  2020  and/or to insert known symbols into each block. The purpose of synchronization code  2032  is discussed in section  3 .  
         [0164]      FIG. 21 , on the other hand, illustrates an example embodiment of a TDM/FDM block  2100  configured for FDM, which comprises sub-block repeater  2102 , sub-block scrambler  2104 , block coder  2106 , sub-block transformer  2108 , sub-block terminator  2110 , and SYNC inserter  2112 , Sub-block repeater  2102  repeats block  2114  and generates block  2116 . Sub-block scrambler then scrambles block  2116  generating block  2118 . Sub-block coder  2106  takes block  2118  and codes it, generating block  2120 . Coding block correlates the data symbols together and generates symbols b. This requires joint demodulation in the receiver, which is more robust but also more complex. Sub-block transformer  2108  then performs a transformation on block  2120 , generating block  2122 . Preferably, the transformation is an IFFT of block  2120 , which allows for more efficient equalizers to be used in the receiver. Next, sub-block terminator  2110  terminates block  2122 , generating block  2124  and SYNC inserter  2112  periodically inserts a synchronization code  2126  after a certain number of blocks  2124  and/or insert known symbols into each block. Preferably, sub-block terminator  2110  only uses cyclic prefix termination as described above. Again this allows for more efficient receiver designs.  
         [0165]     TDM/FDM block  2100  is provided to illustrate the logical components that can be included in a TDM/FDM block configured to perform FDM on a data stream. Depending on the actual implementation, some of the logical components may or may not be included. Moreover, TDM/FDM block  2000  and  2100  are preferably programmable so that the appropriate logical components can be included as required by a particular implementation. This allows a device that incorporates one of blocks  2000  or  2100  to move between different systems with different requirements. Further, it is preferable that TDM/FDM block  1608  in  FIG. 16  be programmable so that it can be programmed to perform TDM, such as described in conjunction with block  2000 , or FDM, such as described in conjunction with block  2100 , as required by a particular communication system.  
         [0166]     After TDM/FDM blocks  1608 , in  FIG. 16 , the parallel data streams are preferably passed to interpolators  1610 .  
         [0167]     After interpolators  1610 , the parallel data streams are passed to filters  1612 , which apply the pulse snapping described in conjunction with the roll-off factor of equation (2) in section  1 . Then the parallel data streams are sent to frequency shifter  1614 , which is configured to shift each parallel data stream by the frequency offset associated with the sub-channel to which the particular parallel data stream is associated.  
         [0168]      FIG. 22  illustrates an example embodiment of a frequency shifter  2200  in accordance with the systems and methods described herein. As can be seen, frequency shifter  2200  comprises multipliers  2202  configured to multiply each parallel data stream by the appropriate exponential to achieve the required frequency shift. Each exponential is of the form: exp(j2πf c nT/rM), where c is the corresponding sub-channel, e.g., c=0 to N−1, and n is time. Preferably, frequency shifter  1614  in  FIG. 5  is programmable so that various channel/sub-channel configurations can be accommodated for various different systems. Alternatively, an IFFT block can replace shifter  1614  and filtering can be done after the IFFT block. This type of implementation can be more efficient depending on the implementation.  
         [0169]     After the parallel data streams are shifted, they are summed, e.g., in summer  1512  of  FIG. 15 . The summed data stream is then transmitted using the entire bandwidth B of the communication channel being used. But the transmitted data stream also comprises each of the parallel data streams shifted in frequency such that they occupy the appropriate sub-channel. Thus, each sub-channel may be assigned to one user, or each sub-channel may carry a data stream intended for different users. The assignment of sub-channels is described in section  3   b . Regardless of how the sub-channels are assigned, however, each user will receive the entire bandwidth, comprising-all the sub-channels, but will only decode those sub-channels assigned to the user.  
         [heading-0170]     6. Sample Receiver Embodiments  
         [0171]      FIG. 23  illustrates an example embodiment of a receiver  2300  that can be configured in accordance with the present invention. Receiver  2300  comprises an antenna  2302  configured to receive a message transmitted by a transmitter, such as transmitter  1500 . Thus, antenna  2302  is configured to receive a wide band message comprising the entire bandwidth B of a wide band channel that is divided into sub-channels of bandwidth B. As described above, the wide band message comprises a plurality of messages each encoded onto each of a corresponding sub-channel. All of the sub-channels may or may not be assigned to a device that includes receiver  2300 ; therefore, receiver  2300  may or may not be required to decode all of the sub-channels.  
         [0172]     After the message is received by antenna  2300 , it is sent to radio receiver  2304 , which is configured to remove the carrier associated with the wide band communication channel and extract a baseband signal comprising the data stream transmitted by the transmitter. Example radio receiver embodiments are described in more detail below.  
         [0173]     The baseband signal is then sent to correlator  2306  and demodulator  2308 , Correlator  2306  is configured to correlated with a synchronization code inserted in the data stream as described in section  3 . It is also preferably configured to perform SIR and multipath estimations as described in section  3 ( b ). Demodulator  2308  is configured to extract the parallel data streams from each sub-channel assigned to the device comprising receiver  2300  and to generate a single data stream therefrom.  
         [0174]      FIG. 24  illustrates an example embodiment of a demodulator  2400  in accordance with the systems and methods described herein. Demodulator  2402  comprises a frequency shifter  2402 , which is configured to apply a frequency offset to the baseband data stream so that parallel data streams comprising the baseband data stream can be independently processed in receiver  2400 . Thus, the output of frequency shifter  2402  is a plurality of parallel data streams, which are then preferably filtered by filters  2404 . Filters  2404  apply a filter to each parallel data stream that corresponds to the pulse shape applied in the transmitter, e.g., transmitter  1500 . Alternatively, an IFFT block can replace shifter  1614  and filtering can be done after the IFFT block. This type of implementation can be more efficient depending on the implementation.  
         [0175]     Next, receiver  2400  preferably includes decimators  2406  configured to decimate the data rate of the parallel bit streams. Sampling at higher rates helps to ensure accurate recreation of the data. But the higher the data rate, the larger and more complex equalizer  2408  becomes. Thus, the sampling rate, and therefore the number of samples, can be reduced by decimators  2406  to an adequate level that allows for a smaller and less costly equalizer  2408 .  
         [0176]     Equalizer  2408  is configured to reduce the effects of multipath in receiver  2300 . Its operation will be discussed more fully below. After equalizer  2408 , the parallel data streams are sent to de-scrambler, decoder, and de-interleaver  2410 , which perform the opposite operations of scrambler, encoder, and interleaver  1506  so as to reproduce the original data generated in the transmitter. The parallel data streams are then sent to parallel to serial converter  2412 , which generates a single serial data stream from the parallel data streams.  
         [0177]     Equalizer  2408  uses the multipath estimates provided by correlator  2306  to equalize the effects of multipath in receiver  2300 . In one embodiment, equalizer  2408  comprises Single-In Single-Out (SISO) equalizers operating on each parallel data stream in demodulator  2400 . In this case, each SISO equalizer comprising equalizer  2408  receives a single input and generates a single equalized output. Alternatively, each equalizer can be a Multiple-In Multiple-Out (MIMO) or a Multiple-In Single-Out (MISO) equalizer. Multiple inputs can be required for example, when a frequency encoder or rate controller, such as frequency encoder  1900 , is included in the transmitter. Because frequency encoder  1900  encodes information from more than one parallel data stream onto each sub-channel, each equalizers comprising equalizer  2408  need to equalize more than one sub-channel Thus, for example, if a parallel data stream in demodulator  2400  comprises d( 1 )+d( 8 ), then equalizer  2408  will need to equalize both d( 1 ) and d( 8 ) together. Equalizer  2408  can then generate a single output corresponding to d( 1 ) or d( 8 ) (MISO) or it can generate both d( 1 ) and d( 8 ) (MIMO).  
         [0178]     Equalizer  2408  can also be a time domain equalizer (TDE) or a frequency domain equalizer (FDE) depending on the embodiment. Generally, equalizer  2408  is a TDE if the modulator in the transmitter performs TDM on the parallel data streams, and a FDE if the modulator performs FDM. But equalizer  2408  can be an FDE even if TDM is used in the transmitter. Therefore, the preferred equalizer type should be taken into consideration when deciding what type of block termination to use in the transmitter. Because of power requirements, it is often preferable to use FDM on the forward link and TDM on the reverse link in a wireless communication system.  
         [0179]     As with transmitter  1500 , the various components comprising demodulator  2400  are preferably programmable, so that a single device can operate in a plurality of different systems and still maintain superior performance, which is a primary advantage of the systems and methods described herein. Accordingly, the above discussion provides systems and methods for implementing a channel access protocol that allows the transmitter and receiver hardware to be reprogrammed slightly depending on the communication system.  
         [0180]     Thus, when a device moves from one system to another, it preferably reconfigures the hardware, i.e. transmitter and receiver, as required and switches to a protocol stack corresponding to the new system. An important part of reconfiguring the receiver is reconfiguring, or programming, the equalizer because multipath is a main problem for each type of system. The multipath, however, varies depending on the type of system, which previously has meant that a different equalizer is required for different types of communication systems. The channel access protocol described in the preceding sections, however, allows for equalizers to be used that need only be reconfigured slightly for operation in various systems.  
         [0181]     a. Sample Equalizer Embodiment  
         [0182]      FIG. 25  illustrates an example embodiment of a receiver  2500  illustrating one way to configure equalizers  2506  in accordance with the systems and methods described herein. Before discussing the configuration of receiver  2500 , it should be noted that one way to configure equalizers  2506  is to simply include one equalizer per channel (for the systems and methods described herein, a channel is the equivalent of a sub-channel as described above), A correlator, such as correlator  2306  ( FIG. 23 ), can then provide equalizers  2506  with an estimate of the number, amplitude, and phase of any multipaths present, up to some maximum number. This is also known as the Channel Impulse Response (CIR). The maximum number of multipaths is determined based on design criteria for a particular implementation. The more multipaths included in the CIR the more path diversity the receiver has and the more robust communication in the system will be. Path diversity is discussed a little more fully below.  
         [0183]     If there is one equalizer  2506  per channel, the CIR is preferably provided directly to equalizers  2506  from the correlator (not shown). If such a correlator configuration is used, then equalizers  2506  can be run at a slow rate, but the overall equalization process is relatively fast. For systems with a relatively small number of channels, such a configuration is therefore preferable. The problem, however, is that there is large variances in the number of channels used in different types of communication systems. For example, an outdoor system can have has many as 256 channels. This would require 256 equalizers  2506 , which would make the receiver design too complex and costly. Thus, for systems with a lot of channels, the configuration illustrated in  FIG. 25  is preferable. In receiver  2500 , multiple channels share each equalizer  2506 . For example, each equalizer can be shared by 4 channels, e.g., CH 1 -Ch 4 , Ch 5 -CH 8 , etc., as illustrated in  FIG. 25 , In which case, receiver  2500  preferably comprises a memory  2502  configured to store information arriving on each channel.  
         [0184]     Memory  2502  is preferably divided into sub-sections  2504 , which are each configured to store information for a particular subset of channels. Information for each channel in each subset is then alternately sent to the appropriate equalizer  2506 , which equalizes the information based on the CIR provided for that channel. In this case, each equalizer must run much faster than it would if there was simply one equalizer per channel. For example, equalizers  2506  would need to run 4 or more times as fast in order to effectively equalize 4 channels as opposed to 1. In addition, extra memory  2502  is required to buffer the channel information. But overall, the complexity of receiver  2500  is reduced, because there are fewer equalizers. This should also lower the overall cost to implement receiver  2500 .  
         [0185]     Preferably, memory  2502  and the number of channels that are sent to a particular equalizer is programmable. In this way, receiver  2500  can be reconfigured for the most optimum operation for a given system. Thus, if receiver  2500  were moved from an outdoor system to an indoor system with fewer channels, then receiver  2500  can preferably be reconfigured so that there are fewer, even as few as 1, channel per equalizer. The rate at which equalizers  2506  are run is also preferably programmable such that equalizers  2506  can be run at the optimum rate for the number of channels being equalized.  
         [0186]     In addition, if each equalizer  2506  is equalizing multiple channels, then the CIR for those multiple paths must alternately be provided to each equalizer  2506 . Preferably, therefore, a memory (not shown) is also included to buffer the CIR information for each channel. The appropriate CIR information is then sent to each equalizer from the CIR memory (not shown) when the corresponding channel information is being equalized. The CIR memory (not shown) is also preferably programmable to ensure optimum operation regardless of what type of system receiver  2500  is operating in.  
         [0187]     Returning to the issue of path diversity, the number of paths used by equalizers  2506  must account for the delay spread d s  in the system. For example, if the system is an outdoor system operating in the 5 gigahertz (GHz) range, the communication channel can comprise a bandwidth of 125 megahertz (MHz), e.g., the channel can extend from 5.725 GHz to 5.85 GHz. If the channel is divided into 512 sub-channels with a roll-off factor r of 0.125, then each sub-channel will have a bandwidth of approximately 215 kilohertz (KHz), which provides approximately a 4.6 microsecond symbol duration. Since the worst case delay spread d s  is 20 microseconds, the number of paths used by equalizers  2504  can be set to a maximum of 5. Thus, there would be a first path P 1  at zero microseconds, a second path P 2  at 4.6 microseconds, a third path P 3  at 9.2 microseconds, a fourth path P 4  at 13.8 microseconds, and fifth path P 5  at 18.4 microseconds, which is close to the delay spread d s  In another embodiment, a sixth path can be included so as to completely cover the delay spread d s ; however, 20 microseconds is the worst case. In fact, a delay spread d s  of 3 microseconds is a more typical value. In most instances, therefore, the delay spread d s  will actually be shorter and an extra path is not needed. Alternatively, fewer sub-channels can be used, thus providing a larger symbol duration, instead of using an extra path. But again, this would typically not be needed.  
         [0188]     As explained above, equalizers  2506  are preferably configurable so that they can be reconfigured for various communication systems. Thus, for example, the number of paths used must be sufficient regardless of the type of communication system. But this is also dependent or the number of sub-channels used. If, for example, receiver  2500  went from operating in the above described outdoor system to an indoor system, where the delay spread d s  is on the order of 1 microseconds, then receiver  2500  can preferably be reconfigured for 32 sub-channels and 5 paths. Assuming the same overall bandwidth of 125 MHz, the bandwidth of each sub-channel is approximately 4 MHz and the symbol duration is approximately 250 ns.  
         [0189]     Therefore, there will be a first path P 1  at zero microseconds and subsequent paths P 2  to P 5  at 250 ns, 500 ns, 750 ns, and 1 microsecond, respectively. Thus, the delay spread d s  should be covered for the indoor environment. Again, the 1 microsecond delay spread d s  is worst case so the 1 microsecond delay spread d s  provided in the above example will often be more than is actually required. This is preferable, however, for indoor systems, because it can allow operation to extend outside of the inside environment, e.g., just outside the building in which the inside environment operates. For campus style environments, where a user is likely to be traveling between buildings, this can be advantageous.  
         [heading-0190]     7 Sample Embodiment of a Wireless Communication Device  
         [0191]      FIG. 26  illustrates an example embodiment of a wireless communication device in accordance with the systems and methods described herein. Device  2600  is, for example, a portable communication device configured for operation in a plurality of indoor and outdoor communication systems. Thus, device  2600  comprises an antenna  2602  for transmitting and receiving wireless communication signals over a wireless communication channel  2618 . Duplexor  2604 , or switch, can be included so that transmitter  2606  and receiver  2608  can both use antenna  2602 , while being isolated from each other. Duplexors, or switches used for this purpose, are well known and will not be explained herein.  
         [0192]     Transmitter  2606  is a configurable transmitter configured to. implement the channel access protocol described above. Thus, transmitter  2606  is capable of transmitting and encoding a wideband communication signal comprising a plurality of sub-channels. Moreover, transmitter  2606  is configured such that the various subcomponents that comprise transmitter  2606  can be reconfigured, or programmed, as described in section  5 . Similarly, receiver  2608  is configured to implement the channel access protocol described above and is, therefore, also configured such that the various sub-components comprising receiver  2608  can be reconfigured,” or reprogrammed, as described in section  6 .  
         [0193]     Transmitter  2606  and receiver  2608  are interfaced with processor  2610 , which can comprise various processing, controller, and/or Digital Signal Processing (DSP) circuits. Processor  2610  controls the operation of device  2600  including encoding signals to be transmitted by transmitter  2606  and decoding signals received by receiver  2608 . Device  2610  can also include memory  2612 , which can be configured to store operating instructions, e.g., firmware/software, used by processor  2610  to control the operation of device  2600 .  
         [0194]     Processor  2610  is also preferably configured to reprogram transmitter  2606  and receiver  2608  via control interfaces  2614  and  2616 , respectively, as required by the wireless communication system in which device  2600  is operating. Thus, for example, device  2600  can be configured to periodically ascertain the availability is a preferred communication system. If the system is detected, then processor  2610  can be configured to load the corresponding operating instruction from memory  2612  and reconfigure transmitter  2606  and receiver  2608  for operation in the preferred system.  
         [0195]     For example, it may preferable for device  2600  to switch to an indoor wireless LAN if it is available. So device  2600  may be operating in a wireless WAN where no wireless LAN is available, while periodically searching for the availability of an appropriate wireless LAN. Once the wireless LAN is detected, processor  2610  will load the operating instructions, e.g., the appropriate protocol stack, for the wireless LAN environment and will reprogram transmitter  2606  and receiver  2608  accordingly, In this manner, device  2600  can move from one type of communication system to another, while maintaining superior performance.  
         [0196]     It should be noted that a base station configured in accordance with the systems and methods herein will operate in a similar manner as device  2600 ; however, because the base station does not move from one type of system to another, there is generally no need to configure processor  2610  to reconfigure transmitter  2606  and receiver  2608  for operation in accordance with the operating instruction for a different type of system. But processor  2610  can still be configured to reconfigure, or reprogram the sub-components of transmitter  2606  and/or receiver  2608  as required by the operating conditions within the system as reported by communication devices in communication with the base station. Moreover, such a base station can be configured in accordance with the systems and methods described herein to implement more than one mode of operation. In which case, controller  2610  can be configured to reprogram transmitter  2606  and receiver  2608  to implement the appropriate mode of operation.  
         [heading-0197]     8 Bandwidth Recovery  
         [0198]     As described above in relation to  FIGS. 11-14 , when a device, such as device  1118  is near the edge of a communication cell  1106 , it may experience interference from base station  1112  of an adjacent communication cell  1104 . In this case, device  1118  will report a low SIR to base station  1114 , which will cause base station  1114  to reduce the number of sub-channels assigned to device  1118 . As explained in relation to  FIGS. 12 and 13 , this reduction can comprise base station  1114  assigning only even sub-channels to device  1118 . Preferably, base station  1112  is correspondingly assigning only odd sub-channels to device  1116 .  
         [0199]     fn this manner, base station  1112  and  1114  perform complementary reductions in the channels assigned to devices  1116  and  1118  in order to prevent interference and improve performance of devices  1116  and  1118 . The reduction in assigned channels reduces the overall bandwidth available to devices  1116  and  1118 . But as described above, a system implementing such a complementary reduction of sub-channels will still maintain a higher bandwidth than conventional systems. Still, it is preferable to recover the unused sub-channels, or unused bandwidth, created by the reduction of sub-channels in response to a low reported SIR.  
         [0200]     One method for recovering the unused bandwidth is illustrated in the flow chart of  FIG. 27 . First, in step  2702 , base station  1114  receives SIR reports for different groups of sub-channels from device  1118  as described above. If the group SIR reports are good, then base station  1114  can assign all sub-channels to device  1118  in step  2704 . If, however, some of the group SIR reports received in step  2702  are poor, then base station  1114  can reduce the number of sub-channels assigned to device  1118 , e.g., by assigning only even sub-channels, in step  2706 . At the same time, base station  1112  is preferably performing a complementary reduction in the sub-channels assigned to device  1116 , e.g., by assigning only odd sub-channels.  
         [0201]     At this point, each base station has unused bandwidth with respect to devices  1116  and  1118 . To recover this bandwidth, base station  1114  can, in step  2708 , assign the unused odd sub-channels to device  1116  in adjacent cell  1104 . It should be noted that even though cells  1102 ,  1104 , and  1106  are illustrated as geometrically shaped, non-overlapping coverage areas, the actual coverage areas do not resemble these shapes. The shapes are essentially fictions used to plan and describe a wireless communication system  1100 . Therefore, base station  1114  can in fact communicate with device  1116 , even though it is in adjacent cell  1104 .  
         [0202]     Once base station  1114  has assigned the odd sub-channels to device  1116 . in step  2708 , base station  1112  and  1114  communicate with device  1116  simultaneously over the odd sub-channels in step  2710 . Preferably, base station  1112  also assigns the unused even sub-channels to device  1118  in order to recover the unused bandwidth in- cell  1104  as well.  
         [0203]     In essence, spatial diversity is achieved by having both base station  1114  and  1112  communicate with device  1116  (and  1118 ) over the same sub-channels. Spatial diversity occurs when the same message is transmitted simultaneously over statistically independent communication paths to the same receiver. The independence of the two paths improves the overall immunity of the system to fading. This is because the two paths will experience different fading effects. Therefore, if the receiver cannot receive the signal over one path due to fading, then it will probably still be able to receive the signal over the other path, because the fading that effected the first path will not effect the second. As a result, spatial diversity improves overall system performance by improving the Bit Error Rate (BER) in the receiver, which effectively increases the deliverable data rate to the receiver, i.e., increase the bandwidth.  
         [0204]     For effective spatial diversity, base stations  1112  and  1114  ideally transmit the same information at the same time over the same sub-channels. As mentioned above, each base station in system  1100  is configured to transmit simultaneously, i.e., system  1100  is a TDM system with synchronized base stations. Base stations  1112  and  1114  also assigned the same sub-channels to device  1116  in step  2708 . Therefore, all that is left is to ensure that base stations  1112  and  1114  send the same information. Accordingly, the information communicated to device  1116  by base stations  1112  and  1114  is preferably coordinated so that the same information is transmitted at the same time. The mechanism for enabling this coordination is discussed more fully below. Such coordination, however, also allows encoding that can provide further performance enhancements within system  1100  and allow a greater percentage of the unused bandwidth to be recovered.  
         [0205]     One example coordinated encoding scheme that can be implemented between base stations  1112  and  1114  with respect to communications with device  1116  is Space-Time-Coding (STC) diversity. STC Is illustrated by system  2800  in  FIG. 28 . In system  2800 , transmitter  2802  transmits a message over channel  2808  to receiver  2806 . Simultaneously, transmitter  2804  transmits a message over channel  2810  to receiver  2806 . Because channels  2808  and  2810  are independent, system  2800  will have spatial diversity with respect to communications from transmitters  2802  and  2804  to receiver  2806 . In addition, however, the data transmitted by each transmitter  2802  and  2804  can be encoded to also provide time diversity. The following equations illustrate the process of encoding and decoding data in a STC system, such as system  2800 .  
         [0206]     First, channel  2808  can be denoted h n  and channel  2810  can be denoted g n , where: 
 
h n =α a e jθh  and  (1) 
 
g n =α h e jθg   (2) 
 
         [0207]     Second, we can look at two blocks of data  2812   a  and  2812   b  to be transmitted by transmitter  2802  as illustrated in  FIG. 28 . Block  2812   a  comprises N-symbols denoted as a 0 , a 1 , a 2 , . . . , a N−1 , or a(0:N−1). Block  2812   b  transmits N-symbols of data denoted b(0:N−1). Transmitter  2804  simultaneously transmits two block of data  2814   a  and  2814   b . Block  2814   a  is the negative inverse conjugate of block  2812   b  and can therefore be described as −b*(N−1:0). Block  2814   b  is the inverse conjugate of block  2812   a  and can therefore be described as a*(N−1:0). It should be noted that each block of data in the forgoing description will preferably comprise a cyclical prefix as described above.  
         [0208]     When blocks  2812   a ,  2812   b ,  2814   a , and  2814   b  are received in receiver  2806 , they are combined and decoded in the following manner: First, the blocks will be combined in the receiver to form the following blocks, after discarding the cyclical prefix: 
 
Block 1 = a (0: N− 1){circle over (×)} h   n   −b *( N− 1:0){circle over (×)}g n ; and  (3) 
 
Block 2 = b (0 :N− 1){circle over (×)}h n   +a *( N− 1:0){circle over (×)}g n   (4) 
        Where the symbol {circle over (×)} represents a cyclic convolution.        
 
         [0210]     Second, by taking an IFFT of the blocks, the blocks can be described as: 
 
Block 1 = A   n*   H   n   −B   n * *   G   n ; and  (5) 
 
Block 2 = B   n*   H   n   −A   n * *   G   n .  (6) 
        Where n=0 to N−1.        
 
         [0212]     In equations (5) and (6) H n  and G n  will be known, or can be estimated. But to solve the two equations and determine A n  and B n , it is preferable to turn equations (5) and (6) into two equations with two unknowns. This can be achieved using estimated signals X, and Y, as follows: 
 
 X   n   =A   n*   H   n   −B   n * * G n;  and  (7) 
 
 Y   n   =B   n* H n   +A   n * * G n   (8) 
 
         [0213]     To generate two equations and two unknowns, the conjugate of Y, can be used to generate the following two equations: 
 
 X   n   =A   n*   H   n   −B   n * * G n ; and  (9) 
 
 Y   n   *=B   n   *   * H n   *+A   n*   G   n *.  (10) 
 
         [0214]     Thus, the two unknowns are A n  and B n * and equations (9) and (10) define a matrix relationship in terms of these two unknowns as follows:  
                 [           X   n               Y   n   *           ]     =       [             H   n     -     G   n                   G   n   *     ⁢     H   n   *             ]     *     [           A   n               B   n   *           ]         ⁢     
             (   11   )                       [           A   n               B   n   *           ]     =       1       H   n     ⁢     |   2     ⁢     +     |     G   n     ⁢     |   2             *     [             H   n   *     ⁢     G   n                   -     G   n   *       ⁢     H   n             ]     *     [           X   n               Y   n   *           ]                                 (   12   )             
 
         [0215]     Signals A n  and B n  can be determined using equation (12). It should be noted, that the process just described is not the only way to implement STC. Other methods can also be implemented in accordance with the systems and methods described herein. Importantly, however, by adding time diversity, such as described in the preceding equations, to the space diversity already achieved by using base stations  1112  and  1114  to communicate with device  1116  simultaneously, the BER can be reduced even further to recover even more bandwidth.  
         [0216]     An example transmitter  2900  configured to communicate using STC in accordance with the systems and methods described herein is illustrated in  FIG. 29 . Transmitter  2900  includes a block storage device  2902 , a serial-to-parallel converter  2904 , encoder  2906 , and antenna  2908 . Block storage device  2902  is included in transmitter  2900  because a 1 block delay is necessary to implement the coding illustrated in  FIG. 28 . This is because transmitter  2804  first transmits −b n * (n=N−1 to 0). But b n  is the second block, so if transmitter  2900  is going to transmit −b n * first, it must store two blocks, e.g., a n  and b n , and then generate block  2814   a  and  2814   b  (see  FIG. 28 ).  
         [0217]     Serial-to-parallel converter  2904  generates parallel bit streams from the bits of blocks a n  and b n . Encoder  2906  then encodes the bit streams as required, e.g., encoder  2906  can generate −b n * and a n * (see blocks  2814   a  and  2814   b  in  FIG. 28 ). The encoded blocks are then combined into a single transmit signal as described above and transmitted via antenna  2908 .  
         [0218]     Transmitter  2900  preferably uses TDM to transmit messages to receiver  2806 . An alternative transmitter  3000  embodiment that uses FDM is illustrated in  FIG. 30 . Transmitter  3000  also includes block storage device  3002 , a serial-to-parallel converter  3004 , encoder  3006 , and antenna  3008 , which are configured to perform in the same manner as the corresponding components in transmitter  2900 . But in addition, transmitter  3000  includes IFFTs  3010  to take the IFFT of the blocks generated by encoder  2906 . Thus, transmitter  3000  transmits −B n * and A n * as opposed to −b n * and a n *, which provides space, frequency, and time diversity.  
         [0219]      FIG. 31  illustrates an alternative system  3100  that also uses FDM but that eliminates the 1 block delay associated with transmitters  2900  and  3000 . In system  3100 , transmitter  3102  transmits over channel  3112  to receiver  3116 . Transmitter  3106  transmits over channel  3114  to receiver  3116 . As with transmitters  2802  and  2804 , transmitters  3102  and  3106  implement an encoding scheme designed to recover bandwidth in system  3100 . In system  3100 , however, the coordinated encoding occurs at the symbol level instead of the block level.  
         [0220]     Thus, for example, transmitter  3102  can transmit block  3104  comprising symbols a 0 , a 1 , a 2 , and a 3 . In which case, transmitter  3106  will transmit a block  3108  comprising symbols −a 1  *, a 0 *, −a 3 *, and a 2 *. As can be seen, this is the same encoding scheme used by transmitters  2802  and  2804 , but implemented at the symbol level instead of the block level. As such, there is no need to delay one block before transmitting. An IFFT of each block  3104  and  3108  can then be taken and transmitted using FDM. An IFFT  3110  of block  3104  is shown in  FIG. 31  for purposes of illustration.  
         [0221]     Channels  3112  and  3114  can be described by H n  and G n , respectively. Thus, in receiver  3116  the following symbols will be formed: 
        (A 0* H 0 )−(A 1 * * G 0 )     (A 1* H 1 )+(A 0 * * G 1 )     (A 2* H 2 )−(A 3 * * G 2 )     (A 3* H 3 )+(A 2 * * G 3 ).        
 
         [0226]     In time, each symbol a n  (n=0 to 3) occupies a slightly different time location. In frequency, each symbol A n  (n=0 to 3) occupies a slightly different frequency. Thus, each symbol An is transmitted over a slightly different channel, i.e., H n  (n=0 to 3) or G n  (n=0 to 3), which results in the combinations above.  
         [0227]     As can be seen, the symbol combinations formed in the receiver are of the same form as equations (5) and (6) and, therefore, can be solved in the same manner, but without the one block delay.  
         [0228]     In order to implement STC or Space Frequency Coding (SPC) diversity as described above, bases stations  1112  and  1114  must be able to coordinate encoding of the symbols that are simultaneously sent to a particular device, such as device  1116  or  1118 . Fortunately, base stations  1112  and  1114  are preferably interfaced with a common network interface server. For example, in a LAN, base stations  1112  and  1114  (which would actually be service access points in the case of a LAN) are interfaced with a common network interface server that connects the LAN to a larger network such as a Public Switched Telephone Network (PSTN). Similarly, in a wireless WAN, base stations  1112  and  1114  are typically interfaced with a common base station control center or mobile switching center. Thus, coordination of the encoding can be enabled via the common connection with the network interface server. Bases station  1112  and  1114  can then be configured to share information through this common connection related to communications with devices at the edge of cells  1104  and  1106 . The sharing of information, in turn, allows time or frequency diversity coding as described above.  
         [0229]     It should be noted that other forms of diversity, such as polarization diversity or delay diversity, can also be combined with the spatial diversity in a communication system designed in accordance with the systems and methods described herein. The goal being to combine alternative forms of diversity with the spatial diversity in order to recover larger amounts of bandwidth. It should also be noted, that the systems and methods described can be applied regardless of he number of base stations, devices, and communication cells involved.  
         [0230]     Briefly, delay diversity can preferably be achieved in accordance with the systems and methods described herein by cyclical shifting the transmitted blocks. For example, one transmitter can transmit a block comprising A 0 , A 1 , A 2 , and A 3  in that order, while the other transmitter transmits the symbols in the following order A 3 , A 0 , A 1 , and A 2 . Therefore, it can be seen that the second transmitter transmits a cyclically shifted version of the block transmitted by the first transmitter. Further, the shifted block can be cyclically shifted by more then one symbol of required by a particular implementation.  
         [heading-0231]     9 Open Loop Architecture  
         [0232]     In a wireless communication system, there is the fundamental problem of how to ensure that a receiver can detect and decode a signal transmitted to it from a transmitter in the system. The transmitted signal will have an associated timing, frequency and phase, all of which can be important to a receiver&#39;s ability to receive the signal. But a receiver often does not have any prior knowledge concerning these characteristics of the transmitted signal, other than a general knowledge of possible ranges. Many conventional wireless communication systems use coherent receiver architectures in order to retrieve that frequency, timing, and phase of received signals.  
         [0233]      FIG. 32  illustrates a conventional coherent system  3200  comprising a transmitter  3202  configured to transmit a signal  3206  via an antenna  3204 , and a receiver  3210  configured to receive signal  3206  via antenna  3208 . In general, transmitter  3202  can be configured to transmit a signal with a center frequency of (f 0 ). But even if receiver  3210  is aware that transmitter  3202  is transmitting at (f 0 ) it may still not be able to effectively receive the signal. This is because there is drift in both the transmitter and receiver, do to fundamental limitations on the accuracy of various components and other losses or errors. Thus, as illustrated in  FIG. 32 , transmitter  3202  can actually be transmitting at (f 0 +δ 1 ), while receiver  3210  is actually receiving at (f 0 +δ 2 ). In this example, δ 1  and δ 2  represent the corresponding drift components. Accordingly, receiver  3210  will require some form of Automatic Frequency Correction (AFC) circuitry  3212  configured to ensure that δ 1  and δ 2  are equal, or sufficiently equal, to allow receiver  3210  to receive signal  3206  correctly. Often AFC is carried out in a digital portion of receiver  3210  and thus requires some form of a Digital to Analog Converter (DAC).  
         [0234]     As illustrated by the waveforms in the bottom half of  FIG. 32 , if δ 1  and δ 2  are not equal, then pulses in received signal  3216 , received by receiver  3210 , can be shorter or longer than the corresponding pulse in transmit signal  3214 , transmitted by transmitter  3202 , as illustrated by the variable portion  3218  of signal  3216 . As a result, receiver  3210  will eventually see an extra pulse or will miss one of the pulses in transmit signal  3214 , causing errors in the decoding process.  
         [0235]     When the frequency is corrected and receiver  3210  is looking at the correct frequency for signal  3206 , receiver  3210  is said to be locked to the correct frequency. Once receiver  3210  is locked, it can begin sampling the received signal to detect the data contained therein. But there can be still another issue: As illustrated in  FIG. 33 , it is important to sample each pulse, e.g., of received signal  3216 , at or near the peak, or maximum, to ensure that there is enough energy to correctly decode the data contained therein. If a pulse is sampled to far from the maximum, then errors are more likely to occur. Thus, a conventional coherent system must take measures to determine the correct timing of receiver  3210  relative to transmitter  3202 .  
         [0236]     A final issue for a conventional coherent system is phase. A coherent receiver must also be capable of recovering the correct phase of transmit signal  3206 .  
         [0237]     A common method for obtaining the correct frequency, timing, and phase information is to use a closed loop architecture. In a closed loop architecture, transmitter  3202  transmits information concerning the correct frequency, timing and phase. Receiver  3210 , makes some gross estimates of the correct frequency, timing, and/or phase sufficient enough to enable it to receive the more granular information from transmitter  3202 , which it can then use for more finite adjustments.  
         [0238]     Thus, transmitter  3202  can transmit frequency information that can be used, e.g., by AFC circuitry  3212 , for frequency adjustment as well as timing and phase information that can be used, e.g., by Phase Lock Loop (PLL) circuitry (not shown) in receiver  3210  to recover the correct timing and phase. It can be preferable, however, to avoid the need for such recovery, and the requisite overhead that it requires. The alternative to a closed loop architecture is an open loop architecture in which receiver  3210  is capable of detecting and decoding signal  3206  without any assistance from transmitter  3202 , One way to achieve this in accordance with the systems and methods described herein, is to insert preamble packets into the data being transmitted from transmitter  3202  to receiver  3210 . First, however, it should be noted that the need for phase information can be eliminated by using a non-coherent receiver  3210 .  
         [0239]     The frame structure illustrated in  FIG. 34  can then be used to ensure that receiver  3210  has accurate enough frequency and timing information to allow it to detect and decode signal  3206 . As can be seen, the data being transmitted by transmitter  3202  can be structured such that it comprises a SYNC packet  3402 , followed by data packets  3404  interspersed with preamble packets  3406 . SYNC packet  3402  can allow receiver  3210  to detect and lock onto signal  3206  as described above. Receiver  3210  can then begin sampling the received information to recover the data contained therein. But as illustrated by waveform  3406  in  FIG. 34 , without the correct timing and frequency information the samples will start to drift away from the maximum, leading to errors.  
         [0240]     Preambles  3406 , however, can act as re-synchronizing packets that allow receiver  3210  to maintain sampling at the maximum as illustrated by waveform  3408 . As can be seen, after SYNC packet  3402 , receiver  3210  is sampling at or near the maximum. But, as before, the samples begin to drift away from the maximum. The subsequent preamble re-synchronizes receiver  3210 , however, so that it is again sampling at or near the maximum. If preambles  3206  are positioned close enough, then noticeable errors due to the drift that inherently occurs can be avoided. In other words, the length of packets  3404  can be designed such that the drift over one packet length is minimized, maintaining any resulting degradation within acceptable limits. For example, a packet length of T/5 or T/8 can be used to limit degradation.  
         [0241]     As described above SYNC packet  3402  can comprise a perfect sequence, or code, as can preambles  3406 . In certain embodiments, SYNC packet  3402  can comprise the same perfect sequence as that used to generate preambles  3406 .  FIG. 35  illustrates one possible configuration for SYNC packet  3402 . Here, the same perfect sequence, or code, (c) is repeated a number of times, e.g., 3, or 4, as illustrated in  FIG. 35 . These repeated codes can then be followed by an inverted version of code (c). Use of an inverted code can be helpful to ensure that the packet is following, once the receiver is locked. For example, as SYNC packet  3402  is correlated, it will result in the peaks illustrated by waveform  3504 . Several copies of code (c) are used to ensure that receiver  3210  correlates SYNC packet  3402  and locks onto the correct frequency, because interference caused, e.g., by fading can prevent receiver  3210  from detecting one or more of the peaks. Thus, for example, if receiver  3210  only detected one or two of the peaks corresponding to code (c), receiver  3210  would have no way of knowing if another code (c) should follow or if data packet  3404  should be next. But, in an embodiment using a SYNC packet structure similar to that illustrated by SYNC packet  3402 , once one or two peaks are detected, receiver  3210  can be configured to then start looking for an inverted peak corresponding to the inverse of code (c). Once this inverted peak is detected, then receiver  3210  will know that data packet  3404  is to follow.  
         [0242]     Specific implementations of code (c) are described in more detail below.  
         [0243]     It should be noted that in a multipath environment, it can be beneficial to actually open up a correlation window in which receiver  3210  is configured to attempt to correlate each preamble at multiple discrete intervals, i.e., the correlator used by receiver  3210  can actually comprise several correlators  3610 ,  3612 , and  3614 , as illustrated in  FIG. 36 , the outputs of which are summed by a summer  3616 . Thus, once SYNC packet  3402  is correlated, receiver  3210  should have an idea of the delay spread (d s ) involved, Receiver  3210  should also know approximately where to look for the peaks of preambles  3206 , Thus, a window can be opened around each preamble peak  3604 ,  3606  so that useful multipaths can also be correlated. In one particular implementation, 3 correlators  3610 ,  3612 , and  3614  can be used, and the window around each peak  3604  and  3606  can be set equal to 2*d s , which for an Ultra Wideband (UWB) application, for example, would be approximately 200 ns maximum.  
         [0244]     Further, because receiver  3210  will know where to search for preamble peaks  3604  and  3606 , full correlators are not necessarily required, saving further resources.  
         [0245]     The systems and methods described in the previous paragraphs can also be used to allow devices configured for different air interface standards to operate in the same environment, without interfering with each other. For example, in UWB applications, there are several competing technologies that use different air interface and communication protocols, but that will conceivably be operating in the same physical areas and within the same frequency bands. The description below describes how interference between devices implementing these different technologies can be eliminated, and how the devices can even be allowed to communicate with the same Service Access Points (SAPs).  
         [0246]      FIG. 37  illustrated two possible competing technologies, e.g., competing UWB technologies, that can be operating in the same environment, thus potentially interfering with each other. One technology can, for example, be implementing Orthogonal Frequency Division Multiplexing (OFDM) as illustrated in the upper half of  FIG. 37 . The other technology can be implementing CDMA. As can be seen, such an OFDM system can comprise three frequency bands, each with a bandwidth B 1 , B 2 , B 3 , respectively, which will normally be equal to each other, an overall bandwidth of (B T1 ), and a center frequency of (f c1 ). Such a CDMA signal can comprise an overall-bandwidth of (B T2 ) and a center frequency of (f c2 ). Depending on the various implementations, B T1  can be un-equal to B T2 , although as explained below the closer they are, the better performance can be. Further, f c1  and f c2  do not need to be equal, but often will be fairly close, since the technologies are being implemented in the same frequency band, which is what leads to potential interference.  
         [0247]      FIG. 38  illustrates a wireless communication environment  3800 , such as a LAN environment, in which several communication protocols are operating in the same frequency band, e.g., protocols implementing the channel access techniques briefly described in  FIG. 37 . SAP  3802  is configured to communicate with a plurality of devices, of which devices  3806 ,  3808 , and  3810  are shown for purpose of illustration. One or more of these devices  3806 ,  3808 , and  3810  can be configured as OFDM devices, CDMA devices, and/or as a device configured to implement still another communication protocol.  
         [0248]     In order for SAP  3802  to communicate with each of devices  3806 ,  3808 , and  3810 , there must be a way for devices  3806 ,  3808 , and  3810  to determine if a given communication is intended for the particular device. In one embodiment, this is achieved using a frame structure such as, or similar to, the open loop frame structure described above, In other words, SAP  3802  can be configured to transmit a frame  3804  that includes a SYNC packet  3812  followed by data packets and known re-synchronizing information, such as the preambles described above.  
         [0249]     Packet  3812  can be divided into sub-packets as illustrated in  FIG. 38 . SYNC packet  3812  can comprise packet detection information that can be used to detect frame  3804 , as described above. It can also comprise frame delimiter information used to determine when frame detection has ended and data is to begin, as well as channel estimation information. Again as described above, frame detection information can comprise a series of codes (G) that can be used to lock onto a transmitted signal. Note that the identifier G is used to identify a particular type of code that is described in detail below. The packet detection information can, therefore, be used for automatic control determinations, diversity selection, e.g., antenna diversity selections, DC removal, energy detection, and packet detection as well as other functions and determinations depending on the implementation.  
         [0250]     Frame delimiter information can comprise an inverted version of code (G) as described above. In addition, however, frame delimiter information can also be used for protocol detection. For example, as illustrated in  FIG. 38 , frame delimiter information can comprise several fields each including code (G) or the inverse of code (G). Of course, the first field in this example is always the inverse of code (G). But the subsequent fields can be used to indicate which protocol frame  3804  is using. For example, three field scan be used to specify up to four different protocols, e.g., including CDMA, OFDM, and other protocols such as a Low Data Rate (LDR) protocol for UWB implementations.  
         [0251]     It should also be noted that significant overhead and resource savings can be achieved if the same code is used through out the various sub-packets comprising SYNC packet  3812 . This is because the same circuitry can then be used to decode each sub-packet and perform the corresponding functions, i.e., the same circuitry and/or logic functionality can be used for packet detection, frame delimiting, protocol identification, channel estimation, etc. Similarly, the same code can be used for the preambles used to re-synchronize the receiver as described above, further consolidating the required circuitry.  
         [0252]     Certain other advantages, and resource savings, can be obtained through the use of certain types of codes. For example, if a random code of length N is used, it is well known that a receiver receiving the code requires N−1 adders and associated registers, or memory, which can be very resource intensive. Recognizing this problem, hierarchical codes have been developed to significantly reduce the number of adders and registers required. An even further reduction in resources can be achieved by using Golay (G) codes. A Golay code is characterized by a set of delays. If, for example, a code of length N=128 is used, the Golay code equivalent will require 7 weights and 7 delays and can be implemented using only 13 adders.  
         [0253]     Code (G) can, depending on the embodiment, be a Golay code or a hierarchical Golay code. A hierarchical Golay code is any code that is spread by a Golay code. For example if a Golay code G 64  of length  64  is used and another code A of length  4  for example A=[1 −1 1 −1] is also used, then a hierarchical Golay code (G) can be formed as follows: G=[G 64  −G 64  G 64  −G 64 ].  
         [0254]     Different SAPs, or Piconets, can then use different Golay codes (G) as follows: 
        Option 1 : Each SAP uses a different Golay Code of length  128      Option  2 : Each SAP uses a different Hierarchical Code of length  128  as follows: 
            SAP 1  uses G=[+G 64  +G 64  +G 64  +G 64 ];     SAP 2  uses G=[+G 64  −G 64  +G 64  −G 64 ];     SAP 3  uses G=[+G 64  +G 64  −G 64  −G 64 ]; and     SAP 4  uses G=[+G 64  −G 64  −G 64  +G 64 ].    
               
 
         [0261]     Option  2  can be preferred since the codes of different SAP s are completely orthogonal and generated from the same code G 64 , which allows the same circuitry can be used to lock to all SAPs.  
         [0262]     Thus, by selecting an advantageous code and using that code through out frame  3804 , significant resource savings can be achieved.  
         [0263]     Bandwidth synchronization is also required in order for devices  3806 ,  3808 , and  3810  to be able to receive frame  3804 . In one embodiment, each device can begin using the same bandwidth. For example, assuming that B T2  is greater than or equal to B T1 , and recalling that the center frequency should be the same or nearly the same, then B T1  can originally be used by all devices. It should be noted that B T1  and B T2  do not need to be equal, but the closer they are, the better performance can be.  
         [0264]     Thus, as illustrated in  FIG. 39 , SYNC packet  3812  can be transmitted using B T1 . If the frequency associated with the frequency bands comprising an OFDM system are F 1 , F 2 , and F 3  respectively, then, as illustrated in  FIG. 39 , at time T 1 , the transmitter can transmit at frequency F 1 , at time T 2  the transmitter can transmit at frequency F 2 , and at time T 3  the transmitter can transmit at frequency F 3 . An OFDM device, of course, will have no trouble receiving a SYNC packet  3812  transmit in this manner. Other types of devices, however, will need to compensate for the frequency hoping inherent in an OFDM system.  
         [0265]     For example, a CDMA device can sample the received signal, convert the samples to a digital signal using an Analog to Digital Converter (ADC) ( 3902 ), and then apply some rotation during down conversion, e.g., using down converter  3904 . The resulting signal can then be filtered and passed through a correlator, such as a Golay correlator  3908 . The portion of the packet residing at F 2  should not need any rotation, since both an OFDM device and a CDMA device operating in system  3800  should be using the same, or nearly the same center frequency; however, the portions of the packet at F 1  and F 3  will need to be rotated in order for, e.g., a CDMA device, to receive the packet correctly.  
         [0266]     In the example embodiment described, the chipping rate for a CDMA device should be 3 times the bandwidth or: 3*B, where B=B 1 =B 2 =B 3 . Therefore, while an OFDM receiver will sample the SYNC packet at a rate equal to the bandwidth B, a CDMA receiver will sample at 3*B.  
         [0267]     If SAP  3802  is a OFDM device, then transmitting SYNC packet  3812  as described will not be a problem. But if it is, e.g., a CDMA device, then again some adjustment, e.g., rotation, of the transmit signal will be required. Again, the portion of the signal at F 2  should be fine, but the portions at F 1  and F 3  should be, e.g., rotated. Further, because rotation means that the data will no longer comprise simple 1 s, and −1 s, it will be understood that a DAC will be required to convert the signal to an analog signal after rotation.  
         [0268]     An alternative approach is to simply use F 2  for SYNC packet  3812 . This eliminates the need for rotation, and any required DACs or ADCs.  
         [0269]     Further, in a UWB system, an LDR device can also be configured to receive SYNC packet  3812 . Here, however, reception and any adjustments can be done in analog circuitry, using a non-coherent architecture, in order to keep costs down. Such circuitry is illustrated by the example receiver  4000  of  FIG. 40 . In receiver  4000 , antenna  4002  couples received signals to a filter  4004 , which then passes the signal to an LNA stage  4006 . LNA stage  4006  can then pass the signal onto a subsequent amplification stage  4008 . The output of amplification stage  4008  can then be sent to delay  4010  and combiner  4012 , which can be configured to multiply the output of amplification stage  4008  with the output of delay  4010 . The output of combiner  4012  can then be integrated over the requisite time period by integrator  4014 .  
         [0270]     The integration in the analog domain performed by integrator  4014  allows receiver  4000  to use a sample rate of: [(B*3)/N], where N is the code length.  
         [0271]     Thus, the approach described can be used with either a coherent or non-coherent receiver, which provides added flexibility and capability.  
         [heading-0272]     10. Position Location  
         [0273]     Position location can also be achieved using the systems and methods described herein. For example, in a UWB system, a device  4110  can be in proximity to four SAPs  4102 - 4108  as illustrated in  FIG. 41 . Each SAP can be implementing a SYNC packet as described above. In order to provide position location capability, SAP  4102  can be configured to use a code, such as a Golay code (G), of a certain length N, e.g., N=128. SAP  4104  can also use the same Golay code, but with an additional, e.g., 8 bits appended to the front. The additional bits can, for example, be copied from the last eight bits of the Golay code. SAP  4106  can then append 16 bits and SAP  4108  can append 24 bits as illustrated in  FIG. 41 . The overall length N T  can be maintained equal, by coping codes or bits onto the end of each packet as required.  
         [0274]     With a little additional circuitry, such as that illustrated in  FIG. 42 , device  4110  can lock onto all four SAPs  4102 - 4108  simultaneously. This allows device  4110  to not only determine which SAP has the strongest signal, but because device  4110  will know the relative timing of the signals received from each SAP, it can determine its location relative to each SAP. The position of device  4110  can then be determined using this relative location information.  FIG. 42  is a diagram illustrating example receiver circuitry  4200  that can be used to lock onto a plurality of SAPS in accordance with one embodiment of the systems and methods described herein.  
         [0275]     With reference to  FIGS. 43 and 44 , additional embodiments of the present invention will now be described. The embodiments described below employ ultra-wideband communication technology. Referring to  FIGS. 43 and 44 , ultra-wideband (UWB) communication technology employs discrete pulses of electromagnetic energy that are emitted at, for example, nanosecond or picosecond intervals (generally tens of picoseconds to hundreds of nanoseconds in duration). For this reason, ultra-wideband is often called “impulse radio.” That is, the UWB pulses may be transmitted without modulation onto a sine wave, or a sinusoidal carrier, in contrast with conventional carrier wave communication technology. Thus, UWB generally requires neither an assigned frequency nor a power amplifier.  
         [0276]     Another example of sinusoidal carrier wave communication technology is illustrated in  FIG. 43 . IEEE 802.11a is a wireless local area network (LAN) protocol, which transmits a sinusoidal radio frequency signal at a 5 GHz center frequency, with a radio frequency spread of about 5 MHz. As defined herein, a carrier wave is an electromagnetic wave of a specified frequency and amplitude that is emitted by a radio transmitter in order to carry information. The 802.11 protocol is an example of a carrier wave communication technology. The carrier wave comprises a substantially continuous sinusoidal waveform having a specific narrow radio frequency (5 MHz) that has a duration that may range from seconds to minutes.  
         [0277]     In contrast, an ultra-wideband (UWB) pulse may have a 2.0 GHz center frequency, with a frequency spread of approximately 4 GHz, as shown in  FIG. 44 , which illustrates two typical UWB pulses.  FIG. 44  illustrates that the shorter the UWB pulse in time, the broader the spread of its frequency spectrum. This is because bandwidth is inversely proportional to the time duration of the pulse. A 600-picosecond UWB pulse can have about a 1.8 GHz center frequency, with a frequency spread of approximately 1.6 GHz and a 300-picosecond UWB pulse can have about a 3 GHz center frequency, with a frequency spread of approximately 3.2 GHz. Thus, UWB pulses generally do not operate within a specific frequency, as shown in  FIG. 43 . In addition, either of the pulses shown in  FIG. 44  may be frequency shifted, for example, by using heterodyning, to have essentially the same bandwidth but centered at any desired frequency. And because UWB pulses are spread across an extremely wide frequency range, UWB communication systems allow communications at very high data rates, such as 100 megabits per second or greater.  
         [0278]     Also, because the UWB pulses are spread across an extremely wide frequency range, the power sampled in, for example, a one megahertz bandwidth, is very low. For example, UWB pulses of one nano-second duration and one milliwatt average power (0 dBm) spreads the power over the entire one gigahertz frequency band occupied by the pulse. The resulting power density is thus 1 milliwatt divided by the 1,000 MHz pulse bandwidth, or 0.001 milliwatt per megahertz (−30 dBm/MHz).  
         [0279]     Generally, in the case of wireless communications, a multiplicity of UWB pulses may be transmitted at relatively low power density (milliwatts per megahertz). However, an alternative UWB communication system may transmit at a higher power density. For example, UWB pulses may be transmitted between 30 dBm to −50 dBm.  
         [0280]     Several different methods of ultra-wideband (UWB) communications have been proposed. For wireless UWB communications in the United States, all of these methods must meet the constraints recently established by the Federal Communications Commission (FCC) in their Report and Order issued Apr. 22, 2002 (ET Docket 98-153). Currently, the FCC is allowing limited UWB communications, but as UWB systems are deployed, and additional experience with this new technology is gained, the FCC may expand the use of UWB communication technology. It will be appreciated that the present invention may be applied to current forms of UWB communications, as well as to future variations and/or varieties of UWB communication technology.  
         [0281]     For example, the April 22 Report and Order requires that UWB pulses, or signals occupy greater than 20% fractional bandwidth or 500 megahertz, whichever is smaller. Fractional bandwidth is defined as 2 times the difference between the high and low 10 dB cutoff frequencies divided by the sum of the high and low 10 dB cutoff frequencies. However, these requirements for wireless UWB communications in the United States may change in the future.  
         [0282]     Communication standards committees associated with the International Institute of Electrical and Electronics Engineers (IEEE) are considering a number of ultra-wideband (UWB) wireless communication methods that meet the current constraints established by the FCC. One UWB communication method may transmit UWB pulses that occupy 500 MHz bands within the 7.5 GHz FCC allocation (from 3.1 GHz to 10.6 GHz). In one embodiment of this communication method, UWB pulses have about a 2-nanosecond duration, which corresponds to about a 500 MHz bandwidth. The center frequency of the UWB pulses can be varied to place them wherever desired within the 7.5 GHz allocation. In another embodiment of this communication method, an Inverse Fast Fourier Transform (IFFT) is performed on parallel data to produce 122 carriers, each approximately 4.125 MHz wide. In this embodiment, also known as Orthogonal Frequency Division Multiplexing (OFDM), the resultant UWB pulse, or signal is approximately 506 MHz wide, and has a 242 nanosecond duration. It meets the FCC rules for UWB communications because it is an aggregation of many relatively narrow band carriers rather than because of the duration of each pulse.  
         [0283]     Another UWB communication method being evaluated by the IEEE standards committees comprises transmitting discrete UWB pulses that occupy greater than 500 MHz of frequency spectrum. For example, in one embodiment of this communication method, UWB pulse durations may vary from 2 nanoseconds, which occupies about 500 MHz, to about 133 picoseconds, which occupies about 7.5 GHz of bandwidth. That is, a single UWB pulse may occupy substantially all of the entire allocation for communications (from 3.1 GHz to 10.6 GHz).  
         [0284]     Yet another UWB communication method being evaluated by the IEEE standards committees comprises transmitting a sequence of pulses that may be approximately 0.7 nanoseconds or less in duration, and at a chipping rate of approximately 1.4 giga pulses per second. The pulses are modulated using a Direct-Sequence modulation technique, and is called DS-UWB. Operation in two bands is contemplated, with one band is centered near 4 GHz with a 1.4 GHz wide signal, while the second band is centered near 8 GHz, with a 2.8 GHz wide UWB signal. Operation may occur at either or both of the UWB bands. Data rates between about 28 Megabits/second to as much as 1,320 Megabits/second are contemplated.  
         [0285]     Thus, described above are three different methods of wireless ultra-wideband (UWB) communication. It will be appreciated that the present invention may be employed using any one of the above-described methods, variants of the above methods, or other UWB communication methods yet to be developed.  
         [0286]     Certain features of the present invention may be employed in an ultra-wideband (UWB) communication system. For example, one embodiment of an UWB communication system employs a communication frame that includes a synchronization section, and at least two data sections and at least two preamble sections. An ultra-wideband receiver synchronizes with the ultra-wideband transmitter using the synchronization section, and may also re-synchronize with the transmitter using at least one preamble section. In another embodiment, the length of the data sections are selected to ensure that drift between the ultra-wideband receiver and transmitter will be minimal before re-synchronization.  
         [0287]     The UWB devices, systems and/or methods in the above-described embodiments communicate with each other by transmitting and receiving a plurality of discrete electromagnetic pulses, as opposed to a substantially continuous carrier wave. Each pulse may have a duration that can range between about 10 picoseconds to about 1 microsecond, and a power that may range between about +30 dBm to about −60 dBm, as measured at a single frequency.  
         [0288]     The present invention may be employed in any type of network, be it wireless, wire, or a mix of wire media and wireless components. That is, a network may use both wire media, such as coaxial cable, and wireless devices, such as satellites, or cellular antennas. As defined herein, a network is a group of points or nodes connected by communication paths. The communication paths may use wires or they may be wireless. A network as defined herein can interconnect with other networks and contain sub-networks. A network as defined herein can be characterized in terms of a spatial distance, for example, such as a local area network (LAN), a personal area network (PAN), a metropolitan area network (MAN), a wide area network (WAN), and a wireless personal area network (WPAN), among others. A network as defined herein can also be characterized by the type of data transmission technology used by the network, such as, for example, a Transmission Control Protocol/Internet Protocol (TCP/IP) network, a Systems Network Architecture network, among others. A network as defined herein can also be characterized by whether it carries voice, data, or both kinds of signals. A network as defined herein may also be characterized by users of the network, such as, for example, users of a public switched telephone network (PSTN) or other type of public network, and private networks (such as within a single room or home), among others. A network as defined herein can also be characterized by the usual nature of its connections, for example, a dial-up network, a switched network, a dedicated network, and a non-switched network, among others. A network as defined herein can also be characterized by the types of physical links that it employs, for example, optical fiber, coaxial cable, a mix of both, unshielded twisted pair, and shielded twisted pair, among others.  
         [0289]     The present invention may be employed in any type of wireless network, such as a wireless PAN, LAN, MAN, or WAN. In addition, the present invention may be employed in wire media, as the present invention dramatically increases the bandwidth of conventional networks that employ wire media, such as hybrid fiber-coax cable networks, or CATV networks, yet it can be inexpensively deployed without extensive modification to the existing wire media network.  
         [0290]     Thus, it is seen that systems and methods of ultra-wideband communications are provided. One skilled in the art will appreciate that the present invention can be practiced by other than the above-described embodiments, which are presented in this description for purposes of illustration and not of limitation. The specification and drawings are not intended to limit the exclusionary scope of this patent document. It is noted that various equivalents for the particular embodiments discussed in this description may practice the invention as well. That is, while the present invention has been described in conjunction with specific embodiments, it is evident that many alternatives, modifications, permutations and variations will become apparent to those of ordinary skill in the art in light of the foregoing description. Accordingly, it is intended that the present invention embrace all such alternatives, modifications and variations as fall within the scope of the appended claims. The fact that a product, process or method exhibits differences from one or more of the above-described exemplary embodiments does not mean that the product or process is outside the scope (literal scope and/or other legally-recognized scope) of the following claims.