Abstract:
A non-quasistatic MOS frequency divider circuit uses a phase lock loop configuration including an antenna coil to induce a differential input signal, an antenna resonating capacitor, a rectifier, a voltage controlled ring oscillator, a phase detector and a loop filter. All transistors used are organic MOS devices of PMOS, NMOS or both PMOS and NMOS varieties. The voltage-controlled oscillator includes a multiple delay stage ring oscillator. The phase detector includes transistors connected as sampling switches to sample the individual oscillator stage voltages into the loop filter. The sampling transistors have gates connected to the coil. The loop filter provides a substantially direct current to a loop amplifier and then to the voltage controlled oscillator delay control input. This configuration results in the voltage controlled oscillator frequency being synchronous to—and at a sub-multiple of the antenna signal frequency. The sampling transistor gates are all connected to the coil and thereby become part of the capacitance of the radio frequency parallel resonant network. The transistor gates are then efficiently switched at the rate of the radio frequency signal with no delay relative to the coil voltage. Operation of the phase detector organic transistors is based on non-quasistatic behavior of the transistor. Non-quasistatic operation results in phase detection at a frequency much higher than the quasistatic limit of transistor unity gain bandwidth.

Description:
The present application is related to and claims priority from U.S. Provisional Application Ser. Nos. 60/536,603, filed Jan. 15, 2004, entitled “Circuitry for the Design of RFID Tags in Polymer Electronic Processes”, 60/539,611, filed Jan. 27, 2004, entitled, “RFID Organic Process Methods”, 60/539,612, filed Jan. 27, 2004, entitled, “RFID Organic Circuit Designs”, and 60/539,610, filed Jan. 27, 2004, entitled, “RFID Organic Flexographic Printline Method”. The disclosures of all four provisional applications are herein specifically incorporated in their entirety by this reference. 

   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The invention relates to organic transistors, and, more particularly, to a phase lock loop circuit and frequency division method suitable for use given the performance constraints of organic transistors. 
   2. Description of the Related Art 
   Organic MOS transistors are similar to silicon metal-oxide-semiconductor transistors in operation. The major difference in construction is that the organic MOS transistor utilizes a thin layer of a semiconducting organic polymer film to act as the semiconductor of the device, as opposed to a silicon layer as used in the more typical in-organic silicon MOS device. 
   Referring now to  FIG. 1 , a cross-sectional diagram of a top-gate bottom contact organic MOS transistor  100  is shown. A metallic region  122  is deposited on an insulating substrate  112  forming the gate  122  of the organic MOS device  100 . A thin dielectric region  120  is placed on top of gate region  122  to electrically isolate it from other layers and to act as the MOS gate insulator. Metallic conductors  118  and  116  are formed on the dielectric region  120  above the gate region  122  such that there is a gap  124  between conductors  116  and  118  overlapping gate metal  122 . The gap  124  is known as the channel region of transistor  100 . A thin film of organic semiconducting material  114  is deposited on dielectric region  120  and over at least a portion of metallic conductors  116  and  118 . A voltage applied between the gate  122  and the source  118  modifies the resistance of the organic semiconductor film  114  in gap region  124  in the vicinity of the interface between semiconductor region  124  and dielectric  120 . This is defined as the “field effect”. When another voltage is applied between the source  118  and the drain  116 , a current flows between the drain and source with a value dependent on both the gate-to-source and the drain-to-source voltages. 
   An organic transistor  200  can also be constructed as a top-gate top contact structure as shown in  FIG. 2 . Conductor layer  222  is deposited and patterned on substrate  212 . A dielectric layer  220  is deposited on conductor layer  222 . A thin film of semiconductor material  214  is deposited on top of dielectric layer  220 . A conductive film is deposited and patterned on top of organic semiconductor  2164  to form conductive source and drain regions  216  and  218 , such that there is a gap  224  that overlaps the underlying gate metal layer  224 . The gap  224  is known as the channel region of transistor  200 . Through a field effect, a voltage is applied between gate conductor  222  and source  218  that modifies the resistance of the organic semiconductor  214  in the gap region  224  in the vicinity of the interface between the semiconductor region  214  and the dielectric  220 . When another voltage is applied between source  218  and drain  216 , a current flows between the drain and the source with a value dependence on both the gate-to-source and the drain-to-source voltages. 
   Organic transistor  300  can also be constructed as a top gate structure as shown in  FIG. 3 . A conductive film is deposited and patterned on an insulating substrate  312  to form conductive regions  318  and  316 . One of these conductive regions is known as the source  318 , and the other as the drain  316 . The gap  324  between source  318  and drain  316  is known as the channel region of transistor  300 . A thin organic semiconductor layer  326  is deposited on top of these conductive regions such that the entire gap  324  and at least a portion of conductive regions source  318  and drain  316  are covered. A dielectric layer  320  is deposited on top of semiconductor layer  326 . A conductive layer  322  is deposited and patterned such that at the underlying gap  324  and at least a portion of the source  316  and the drain  316  are covered. A field effect will cause the resistance of the organic semiconductor  326  inside the gap  324  in the vicinity of the interface between the semiconductor  326  and the dielectric  320  to decrease as a voltage is applied between the gate  322  and the source  318 . When another voltage is applied between the source  318  and the drain  316 , current flows between the source  318  and the drain  316 . The value of the current depends on the voltage between gate  322  and the source  318 . 
   In all of the structures shown in  FIGS. 1–3 , all layers may be patterned as long as the gate conductor overlaps the channel region gap and at least a portion of the source and drain, and organic semiconductor and dielectric are placed so that the gate conductor and the source/drain conductor are electrically isolated. 
   Organic semiconductor materials are often classified as polymeric, low molecular weight, or hybrid. Pentacene, hexithiphene, TPD, and PBD are examples of low weight molecules. Polythiophene, parathenylene vinylene, and polyphenylene ethylene are examples of polymeric semiconductors. Polyvinyl carbazole is an example of a hybrid matrial. These materials are not classified as insulators or conductors. Organic semiconductors behave in a manner that can be described in terms analogous to the band theory in inorganic semiconductors. However, the actual mechanics giving rise to charge carriers in organic semiconductors are substantially different from inorganic semiconductors. In inorganic semiconductors, such as silicon, carriers are generated by introducing atoms of different valencies into a host crystal lattice, the quantity of which is described by the number of carriers that are injected into the conduction band, and the motion of which can be described by a wave vector k. In organic semiconductors, carriers are generated in certain materials by the hybridization of carbon molecules in which weakly bonded electrons, called π electrons, become delocalized and travel relatively far distances from the atom which originally gave rise to that electron. This effect is particularly noted in materials comprising of conjugated molecules or benzene ring structures. Because of the delocalization, these π electrons can be loosely described as being in a conduction band. This mechanism gives rise to a low charge mobility, a measure describing the speed with which these carriers can move through the semiconductor, resulting in dramatically lower current characteristics of organic semiconductors in comparison to inorganic semiconductors. 
   Besides a lower mobility, the chemistry of carrier generation gives rise to another key difference between the operation of an organic MOS transistor and inorganic semiconductor. In the typical operation of an inorganic semiconductor, the resistance of the channel region is modified by an “inversion layer” consisting of the charge carriers made up of the type of charge that exists as a minority in the semiconductor. The silicon bulk is doped with the opposite type of carrier as compared to that used for conduction. For example, a p-type inorganic semiconductor built with an n-type semiconductor, but used p-type carriers, also called holes, to conduct current between the source and drain. In the typical operation of an organic semiconductor, however, the resistance of the channel region is modified by an “accumulation layer” consisting of charge carriers made up of the type of charge that exists as a majority in the semiconductor. For example, a PMOS organic transistor uses a P-type semiconductor and p-carriers, or holes, to generate the current in typical operation. 
   To fully understand the operation of a typical organic transistor, “non-quasistatic MOS transistor operation” must be explained. MOS transistors, both organic and inorganic, are normally assumed to allow immediate current flow between the source and drain of the device upon the application of a gate-to-source voltage. This is called the “quasistatic” assumption and allows for the development of a very simple transient-effect model of the MOS device. This assumption is only true when the transistor is operated at a frequency of operation substantially lower than the maximum frequency response of the charge carrier. Whereas this is true for most typical applications using inorganic semiconductors, this is not true for organic transistors operating at high speeds. When there is a significant delay between the application of gate-to-source voltage and the motion of the charge carriers giving rise to current the transient behavior of those charge carriers must be taken into account. 
   This delay has two components: a period with no current flow and a period of increasing current flow until a constant, stable current flow forms. This is shown in  FIG. 4 . The timing diagram of  FIG. 4  includes a gate voltage pulse  424 , a quasistatic drain current pulse  428 , as is found in a conventional silicon MOS transistor, and a “non-quasistatic” drain current pulse  426  as is found in an organic transistor operating at high speed. Referring to voltage pulse  426 , beyond the point where the current no longer increases, the device has quasi-static (“QS”) behavior. The delay region models non-quasistatic (“NQS”) behavior. This region is normally ignored because this delay is typically on the order of picoseconds for silicon MOS circuits that operate with pulse periods of one hundred or more picoseconds. Non-quasistatic behavior can be ignored in this case because the NQS delay is inconsequential relative to the signal periods of interest in a typical silicon MOS circuit. In organic transistors, this delay is on the order of ten nanoseconds, thereby requiring accounting of this effect when the transistor is operated in the hundreds of kilohertz and above range. The unity gain frequency of a transistor is defined as the frequency of operation at which the transistor is has an output voltage equal to the input voltage. When the transistor is operated below this frequency, the output voltage will be larger than the input voltage. When the transistor is operated above this frequency, the gain of the transistor is below unity meaning that the output voltage is less than the input voltage. Unity gain is always well below the frequency at which non-quasistatic behavior becomes an appreciable and measurable effect. 
   Though organic transistors have much lower performance than inorganic transistors, the materials and processing techniques to produce organic transistors cost significantly less those used to produce inorganic transistors. Therefore, organic transistor technology has application where low cost is desired and low performance is acceptable. Therefore, as the effective performance of an organic transistor is increased, the number of applications for organic transistor technology also increase. An example of this type of application are Radio Frequency Identification (RFTD) tags. Though RFID tags can be produced that operate at any frequency, it is desirable to produce RFID tag using frequency ranges that are used in typical applications. One such typical frequency for RFID tags is 13.56 Mhz, a frequency that is well above the unity gain frequency of organic transistors, and in the range where non-quasi-static behavior needs to be taken into account. 
   What is desired, therefore, is a practical circuit, such as a phase lock loop, that uses organic transistors operating at frequencies far above the unity gain bandwidth where non-quasi-static behavior needs to be taken into account. 
   SUMMARY OF THE INVENTION 
   According to an embodiment of the present invention, a non-quasistatic MOS frequency divider circuit uses a phase lock loop configuration comprising an antenna coil to induce a differential input signal, an antenna resonating capacitor, a rectifier, a voltage controlled ring oscillator, a phase detector and a loop filter. The voltage-controlled oscillator is to operate at frequencies below that of the supplied antenna voltage and comprises a series of delay stages with an inverting feedback path. The number of stages in the ring oscillator establish the frequency division ratio. The output of the ring oscillator drives a phase detector comprising one or two organic MOS transistors for each stage of the ring oscillator. The transistors act as sampling switches to sample the output of each stage for use by the loop filter. The signal from the antenna drives a rectifier to develop direct current power and results in the antenna terminal voltages operating each with an average value above the rectifier output negative voltage and below the rectifier output positive voltage. The differential input signal from the antenna is connected to the phase detector transistor gates to control the phase detector sampling operations. Capacitive loading effects of the phase detector transistor gates are absorbed in the capacitance of the antenna tuned-circuit resulting in a large signal voltage at the gates. The gates are operated at frequencies in excess of their associated transistor transition frequency forcing the transistor into the non-quasi-static mode. Each transistor passes a small pulse of current related to the sampled level from the associated delay stage of the ring oscillator. The current pulse is sampled by the loop filter network and is converted into a substantially direct current signal. The signal level is increased by the loop amplifier and returned to the control voltage input of the voltage-controlled oscillator. The control voltage input modifies the delay of each delay stage of the oscillator and thereby modifies the frequency of oscillation. This completes the feedback connection of the phase lock loop. Perturbations in the voltage-controlled oscillator frequency or phase result in a shift of the average output voltage of the loop filter such that the frequency error between the divided antenna signal and voltage-controlled oscillator is corrected. The loop is configured to cause the voltage controlled oscillator to acquire and maintain a frequency that is a synchronous sub-multiple of the antenna signal frequency defined by the number of delay stages in the oscillator. The circuit of the present invention operates as a frequency divider because the phase detector transistors can be switched by the antenna circuit within a channel formation time of less than the 36.9 nanoseconds determined by a half-cycle at the 13.56 MHz radio frequency on the coil. 

   
     DESCRIPTION OF THE DRAWINGS 
     The present invention is illustrated by way of example and not by limitation in the accompanying figures in which like reference numerals indicate similar elements and in which: 
       FIGS. 1–3  are a cross-sectional views of an organic MOS transistors including an insulating substrate, organic polymer film, dielectric layer, and conductive gate, according to the prior art; 
       FIG. 4  is a timing diagram showing a gate voltage pulse, as well as accompanying drain current responses for a quasistatic mode of operation as is found in an ideal silicon MOS transistor, and for a non-quasistatic mode of operation as is found in a typical organic MOS transistor; 
       FIG. 5  is a circuit diagram of a phase lock loop frequency divider circuit according to a first embodiment of the invention including an antenna coil, an antenna resonating capacitor, a PMOS organic transistor phase detector circuit, a voltage-controlled oscillator, a loop filter and a loop amplifier; 
       FIG. 6  is a timing diagram for the circuit of  FIG. 5 , including an antenna input voltage waveform, voltage controlled oscillator stage waveforms, positive and negative loop filter input current waveforms and the loop filter input current difference, operating in the locked mode; 
       FIG. 7  is a timing diagram for the circuit of  FIG. 5 , including an antenna input voltage waveform, voltage controlled oscillator stage waveforms, positive and negative loop filter input current waveforms and the loop filter input current difference, operating in the un-locked mode; 
       FIG. 8  is a circuit diagram of a phase lock loop frequency divider circuit according to a second embodiment of the invention including an antenna coil, an antenna resonating capacitor, a NMOS organic transistor phase detector circuit, a voltage-controlled oscillator, a loop filter and a loop amplifier; 
       FIG. 9  is a circuit diagram of a first type of delay circuit used in the voltage-controlled oscillator of  FIGS. 5 and 8  utilizing PMOS organic transistors; 
       FIG. 10  is a circuit diagram of a second type of delay circuit used in the voltage controlled oscillator of  FIGS. 5 and 8  utilizing PMOS organic transistors; 
       FIG. 11  is a circuit diagram of a third type of delay circuit used in the voltage-controlled oscillator of  FIGS. 5 and 8  utilizing PMOS organic transistors; 
       FIG. 12  is a circuit diagram of a fourth type of delay circuit used in the voltage-controlled oscillator of  FIGS. 5 and 8  utilizing both PMOS and NMOS organic transistors; 
       FIG. 13  is a circuit diagram of a first type of loop filter circuit for use in the circuits of  FIGS. 5 and 8  utilizing PMOS organic transistors; 
       FIG. 14  is a circuit diagram of a second type of loop filter circuit for use in the circuits of  FIGS. 5 and 8  utilizing NMOS organic transistors; and 
       FIG. 15  is a circuit diagram of a third type of loop filter circuit for use in the circuits of  FIGS. 5 and 8  utilizing capacitors and organic transistors. 
   

   DETAILED DESCRIPTION 
   Referring now to  FIG. 5  a phase lock loop frequency divider  500  according to a first embodiment of the present invention includes first and second input terminals for receiving a differential input signal from antenna coil  502  and an output terminal  532  for providing a synchronous output signal, which is generated by voltage controlled oscillator  508 . A capacitor  504  is coupled between the first and second input terminals. The voltage controlled oscillator  508  comprises delay stages  510 ,  512 ,  514 ,  516 ,  518  and  520 , as well as feedback inverter stage  522 . The inverting stage  522  may be included as part of stage  520  depending on the implementation of stage  520 . The delay stages include an delay control input fed by loop amplifier  526 . The voltage controlled oscillator  508  implements a ring oscillator. Each delay stage in the voltage-controlled oscillator provides time delay and phase inversion of the pulse signal passing through the oscillator loop. The feedback inverter  522  guarantees that the total feedback loop within voltage-controlled oscillator  508  has an odd number of inversions, a necessary condition for oscillation of a ring oscillator. The individual stage outputs are connected via phase detector organic PMOS sampling transistors M 1 , M 2 , M 3 , M 4 , M 5  and M 6  to the loop filter  524  through the source connection of the transistors. The gates of transistors M 1 , M 3  and M 5  are connected to the first output of antenna coil  502 . The gates of transistors M 2 , M 4  and M 6  are connected to the second output of antenna coil  502 . The resulting gate interconnection results in alternate voltage controlled oscillator delay stage outputs having sampling transistors connected to the same antenna coil  502  terminal. The drain connection of M 1 , M 3  and M 5  are connected to the first terminal of loop filter  524 . The drain connection of M 2 , M 4  and M 6  are connected to the second terminal of loop filter  524 . The first terminal of loop filter  524  has transistors connected to it that are simultaneously connected to the first terminal of antenna coil  502 . The second terminal of loop filter  524  has transistors connected to it that are simultaneously connected to the second terminal of antenna coil  502 . The loop filter  524  output is connected to loop amplifier  526 . This connection is normally differential to reduce impulse noise sensitivity of the phase lock loop. The output of loop amplifier  526  may be single-ended or differential depending on the implementation details of loop amplifier  526  or voltage controlled oscillator  508 . 
   The length of the ring oscillator may be changed from the six stages shown. If the feedback inverter  522  is included, the total number of delay stages must be even, thus for STAGE N  520 , the value of N must be an even number to support oscillation. If the number of stages is N odd, then the feedback inverter  522  must be removed and the output of delay stage  520  is connected to the input of delay stage  510  to support oscillation. The feedback inverter time delay is less than the time delay of a delay stage. One PMOS phase detector sampling transistor must be connected in the same manner as that shown for M 1 , M 2 , M 3 , M 4 , M 5  and M 6  for every delay stage in the voltage controlled oscillator  508 , regardless of the total value of N. 
   Antenna coil  502  is resonated by the parallel combination of tuning capacitor  504  and the gate capacitances of PMOS transistors M 1 , M 2 , M 3 , M 4 , M 5  and M 6 . Antenna coil  502  senses an alternating magnetic field at the frequency selected for resonance of the loaded antenna circuit comprising antenna coil  502 , tuning capacitor  504 , rectifier  506  and transistors M 1 , M 2 , M 3 , M 4 , M 5  and M 6 . The voltage drive to transistors M 1 , M 2 , M 3 , M 4 , M 5  and M 6  is maximized at resonance and causes the transistors to switch at the signal frequency present at the antenna. The antenna frequency is higher than the transition frequency of the transistors. The transistors, therefore, operate in non-quasi-static mode. The transistors are fully enabled or disabled since the coil voltage is guaranteed to be larger than that of the direct current power supply provided by full-wave rectifier  506 . Full-wave rectifier  506  provides direct current outputs  528  and  530  as the power supply for voltage-controlled oscillator  508  and loop amplifier  526 . The power supply inputs for the delay stages are shown and described in further detail below, and particularly with respect to  FIGS. 9-12 . A second effect of the rectifier  506  is that the antenna coil voltage swings from a voltage less than the negative output  530  of rectifier  506  to a voltage greater than the positive output  528  of rectifier  506 . The negative output  530  is the power supply ground and the positive output  528  is the positive power supply voltage. Voltage-controlled oscillator  508  transistors use this power supply. The gate voltages, therefore, on transistors M 1 , M 2 , M 3 , M 4 , M 5  and M 6  in excess of the power supply values will cause the phase detector sampling transistors to switch with maximum difference in transistor channel charge flow between their enabled and disabled states. 
   The transistors of phase detector of phase lock loop circuit  508  constructed using organic MOS transistors need not have gain at the signal frequency as in a traditional silicon-based circuit. Moreover, the signal drive to the gate of the transistor is in voltage mode from a parallel-tuned inductor-capacitor network. The capacitance of the gate is absorbed into the total capacitance of the tuned network. The result is that the gate voltage can be large depending on the Q, or quality factor of the network. The purpose of the phase lock loop  500  is to cause the frequency of the voltage-controlled oscillator  508  to be a sub-multiple of the input alternating current (AC) signal. 
   Referring to  FIG. 6 , a plot of the timing of the phase lock loop in the locked condition is shown. The input voltage is a sinusoid operating at 13.56 MHz. For purposes of explanation, a divide ratio N of four is chosen. This requires that there are four delay stages in the voltage-controlled oscillator. Only the outputs of the four stages  510 ,  512 ,  514  and  516  are shown. Stages  518  and  520  are not used and the output of stage  516  is connected to the input of feedback inverter  522  for four-stage operation. The vertical dashed lines show the center of the sampling instants for the switches. Alternate groups of switches perform sampling on every other sampling instant. Therefore, on one sampling instant, the positive input of loop filter  524  is driven and on the next sampling instant, the negative input of loop filter  524  is driven. The sampling periods are fairly short due to the non-quasi-static response of the sampling transistors to the peak of the driving sine wave. Due to the weak current drive of the sampling transistors M 1  through M 6  their associated output appears as a current signal. Since two or more switches are enabled at a time on the same signal to the input of loop filter  524 , the currents are summed. Therefore, a if a negative signal occurs at the output of delay stage  510  and a positive signal occurs at the output of delay stage  512  while both switches M 1  and M 3  are enabled, the currents in the switches are in opposing directions and cancel resulting in no current into the loop filter on the associated loop filter  524  input. If a positive signal occurs at the output of delay stage  510  and a positive signal occurs at the output of delay stage  512  while both switches M 1  and M 3  are enabled, the currents in the switches are in aiding directions and the resulting current into the loop filter is positive on the associated loop filter  524  input. 
   In the locked condition the output frequency of the voltage-controlled oscillator is exactly one divided by N, or for the example shown in  FIG. 6 , one-fourth of the antenna input frequency. The first sample shown is that from the output of stage  1 ,  510  and stage  3 ,  514  being transferred via transistors M 1  and M 3  to the positive input of loop filter  524 . The next sampling instant transfers the outputs from stage  2 ,  512  and stage  4   516  via M 2  and M 4  to the negative input of loop filter  524 . The differential nature of the loop filter  524  and loop amplifier  526  performs a subtraction of the two input current signals resulting in the loop filter difference plot at the bottom of  FIG. 6 . The effect on the input of the loop filter is that it receives two current pulses in the negative direction, followed by four current pulses in the positive direction, then two further current pulses in the negative direction. This process repeats indefinitely as long as the antenna and output signal  532  are synchronized at the target frequency ratio. The average of the pulse sequence is zero, indicating that the frequency relationship between the antenna signal and the voltage-controlled oscillator output  532  is correct. The loop filter  524  integrates this signal and maintains the applied voltage to the voltage-controlled oscillator  508  in order to maintain a constant frequency at the output  532 . 
   Referring to  FIG. 7 , a plot of the timing of the phase lock loop in the un-locked condition is shown. The input voltage is a sinusoid operating at 13.56 MHz. For purposes of explanation, a divide ratio N of four is chosen as in the example of  FIG. 6 . 
   In the un-locked condition the output frequency of the voltage-controlled oscillator  508  is not one divided by N, or for this drawing, one-fourth of the antenna input frequency. The first sample shown is that from the output of stage  1 ,  510  and stage  3 ,  514  being transferred via transistors M 1  and M 3  to the positive input of loop filter  524 . The next sampling instant transfers the outputs from stage  2 ,  512  and stage  4 ,  516  via M 2  and M 4  to the negative input of loop filter  524 . The differential nature of the loop filter  524  and loop amplifier  526  performs a subtraction of the two input current signals resulting in the loop filter difference plot at the bottom of  FIG. 7 . Since the frequency relationship between the antenna signal and the voltage-controlled oscillator output  532  is near one-third, the effect on the input of the loop filter is that it receives one current pulses in the negative direction, followed by two current pulses in the positive direction, then one further current pulses in the negative direction. The next two samples are zero. For the first six samples, the average is zero. However, at the seventh sample the output is negative followed by a large positive sample. The ninth and tenth samples are zero, followed by another large positive sample. At this point, the average of the loop filter difference is substantially positive, indicating that the frequency relationship between the antenna signal and the voltage-controlled oscillator output  532  is incorrect. The loop filter integrates the positive average of the samples. The output signal from the filter is amplified in loop amplifier  526  and applied to voltage-controlled oscillator  508  to reduce its output frequency until lock is again achieved. This phase detection scheme senses phase error resulting from increasing or decreasing frequency to result in corrective action. 
   The fastest voltage-controlled oscillator frequencies possible with the present state of the art for the organic MOS transistor voltage-controlled oscillator are on the order of 1 to 2 MHz. For RFID operation, 423.75 kHz is the required voltage-controlled oscillator frequency and is fairly easily achieved with transistor mobilities exceeding 0.1 V-cm 2  at gate lengths near 3 microns. The divide ratio is 13.56 MHz divided by 423.75 kHz resulting in a division ratio N of 32, thereby requiring the use of 32 delay stages. The non-quasistatic delay of the phase detector sampling transistors M 1  through M 6  is on the order of 20 nanoseconds, below the half-period of 36.9 nanoseconds at 13.56 MHz operation. The result is that the phase detector can operate above the required RFID frequency and control an oscillator capable of supporting required RFID data processing. 
   Alternative circuit configurations can be used other than the one shown in  FIG. 5 . Referring now to  FIG. 8  a phase lock loop frequency divider  800  according to a second embodiment of the present invention is virtually identical to that of the first embodiment  500  of  FIG. 5  except that PMOS transistors M 1 , M 2 , M 3 , M 4 , M 5  and M 6  have been replaced with NMOS transistors M 7 , M 8 , M 9 , M 10 , M 11  and M 12 . Operation of phase lock loop  800  is similar to that of phase lock loop  500  except that the NMOS transistors M 7 , M 8 , M 9 , M 10 , M 11  and M 12  are enabled with a positive voltage and disabled with a negative voltage applied to their associated gates. 
   Referring to  FIG. 9 , a first embodiment of a delay circuit  900  to be used in the voltage-controlled oscillators  508  or  808  comprises PMOS transistors M 97  and M 98 . Transistor M 97  has its source connected to the positive power supply, its gate to the delay stage input  902  and its drain to delay stage output  904 . Transistor M 98  has its source connected to delay stage output  904 , its gate to delay control input  906  and its drain connected to power supply ground. A rising input signal  902  causes a falling output signal  904 . The rate of change of the output signal is controlled by the current drawn by M 98  that is set by the voltage on delay input  806 . Similar action occurs for a falling input signal  902  and rising output signal  904 . Variation of the delay control voltage  906  causes a variation of the delay of a pulse signal from input  902  to output  904 . 
   Referring to  FIG. 10 , a second embodiment of a delay circuit  1000  to be used in the voltage-controlled oscillators  508  or  808  comprises PMOS transistors M 105 , M 106 , M 107  and M 108 . Transistor M 105  has its source connected to the positive power supply its gate to the delay stage input  1002  and its drain to the gate of transistor M 108 . Transistor M 106  has its source connected to the gate of transistor M 108 , its gate to delay control input  1006  and its drain connected to power supply ground. Transistor M 108  has its source connected to delay stage output  1004 , its gate to the drain of M 105  and its drain connected to power supply ground. Transistor M 107  has its source connected to positive power supply, its gate connected to a bias voltage  1008  and its drain connected to stage delay output  1004 . Transistors M 105  and M 106  operate as in previously described delay cell  900 . Transistors M 107  and M 108  operate as a voltage level shifter. The level shifter improves signal compatibility between oscillator delay stages by moving the output voltage transition level of the stage to a voltage near the input transition level. The bias voltage  1008  is set to optimize this balance. A rising input signal  1002  causes a falling output signal  1004 . The rate of change of the output signal is controlled by the current drawn by M 106  that is set by the voltage on delay input  1006 . Similar action occurs for a falling input signal  1002  and rising output signal  1004 . Variation of the delay control voltage  1006  will cause a variation of the delay of a pulse signal from input  1002  to output  1004 . 
   Referring to  FIG. 11 , a third embodiment of a delay circuit  1100  to be used in the voltage-controlled oscillators  508  or  808  comprises PMOS transistors M 114 , M 115 , M 116 , M 117  and M 118 . Transistor M 114  has its source connected to the positive power supply, its gate to the delay stage input  1102  and its drain to the sources of transistors M 115  and M 116 . M 115  and M 116  are configured as a differential amplifier with their gates connected to the positive input  1104  and negative input  1106 , respectively and their drains connected to the negative output  1110  and positive output  1108 , respectively. Load transistors M 116  and M 118  have their drains connected to the power supply ground. M 116  and M 118  are constant-current source connected with their gates connected to their associated drains. The drain of M 116  is connected to the negative output  1110  and the drain of M 118  is connected the positive output  1108 . Change in the voltage at the delay control input  1102  causes a change in available current to be switched by M 115  and M 117  between load devices M 116  and M 118 . A decrease in the available current from M 114  results in more time for to charge the input capacitance of the adjacent delay stage connected to the outputs  1108  and  1110  with the effect of increasing the time delay of a signal from inputs  1104  and  1106  to the outputs  1108  and  1110 . 
   Referring to  FIG. 12 , a fourth embodiment of a delay circuit  1200  to be used in the voltage-controlled oscillators  508  or  808  comprises PMOS transistors M 125  and M 126  and NMOS transistors M 127  and M 128 . Transistor M 125  is connected as a current source to the positive power supply with current under control of the voltage at DELAYP  1202 . Transistor M 128  is connected as a current source to the power supply ground with current under control of the voltage at DELAYN  1206 . Transistors M 126  and M 127  comprise a complementary MOS inverter with input  1204  and output  1208 . The inverter is current starved by reduction of current from current sources M 125  and M 128 . Charging time of load capacitances present at output  1208  is increased with a resulting increase of time delay of a signal between the input  1204  and the output  1208 . This method of delay control requires two control signals  1202  and  1206  from the loop amplifier  524  and these are differential in nature. 
   Referring to  FIG. 13 , a first embodiment of a loop filter circuit  1300  to be used in the phase lock loops  500  or  800  comprises PMOS transistors M 132  through M 137 . Transistors M 133 , M 134 , M 136  and M 137  have both source and drain connections to power supply ground. These are operated as capacitors. M 132  and M 135  are operated as biased resistors by connecting their gates to a bias voltage  1304 . M 132  then connects the gate of M 133  to the positive input/output  1302  via its source-drain channel. Similarly, M 135  then connects the gate of M 136  to the negative input/output  1306  via its source-drain channel. The gate of M 134  is connected to the positive input/output  1302 . The gate of M 137  is connected to the positive input/output  1302 . The circuit implements a proportional-plus-integral (PPI) loop filter for use with the phase lock loop. The two inputs receive signals from the two outputs of the sampling phase detector. The phase detector has a finite series resistance due to the relatively poor conduction of the sampling transistors. The phase detector resistance is in series with the resistance of M 132  and M 135 . If M 133  and M 136  were infinitely large capacitors, the two resistances implement a voltage divider such that there is a signal loss from the output of the delay stages to the output of the loop filter. This is the proportional component of the PPI filter. The finite capacitances of M 133  and M 136  interact with M 132  and M 135  to implement a zero in the transfer function of the filter. This is the integral portion of the response. The PPI filter technique is used to provide a stable second-order closed-loop system for the phase lock loop. Other filters are possible, but the PPI approach has the most flexibility and bandwidth. Transistors M 134  and M 137  implement a third pole in the overall phase lock loop closed-loop frequency response. This is used to reduce residual alternating current ripple at the output of the loop filter due to switching of the phase detector transistors. 
   Referring to  FIG. 14 , a second embodiment of a loop filter circuit  1400  to be used in the voltage controlled oscillators  508  or  808  comprises NMOS transistors M 142  to M 147 . This is similar to previously described loop filter circuit  1300 , except the PMOS transistors have been replaced with NMOS transistors. Operation is identical to that of loop filter circuit  1300 , except that bias voltage  1404  must be adjusted to result in resistive operation of M 142  and M 145 . 
   Referring to  FIG. 15 , a third embodiment of a loop filter circuit  1500  to be used in the voltage controlled oscillators  508  or  808  comprises NMOS transistors M 152  and M 155  and capacitors C 153 , C 154 , C 156  and C 157 . This is similar to previously described loop filter circuit  1400 , except the NMOS capacitor connected transistors have been replaced with capacitor elements. Operation is identical to that of loop filter circuit  1400 . 
   While the invention has been described with respect to specific examples including presently preferred modes of carrying out the invention, those skilled in the art will appreciate that there are numerous variations and permutations of the above described systems and techniques that fall within the spirit and scope of the invention as set forth in the appended claims.