Abstract:
A protection circuit for Darlington amplifiers prevents destructive voltage overshoot conditions from occurring. The Darlington amplifier has a pair of transistors connected in a Darlington configuration. A biasing network is coupled to the transistors for supplying a bias voltage to the transistors. A pair of de-coupling capacitors is coupled to the transistors. One of the de-coupling capacitors is coupled to the input of the Darlington amplifier and the other is coupled to the output. The output capacitor has a larger capacitance than the input capacitor such that excessive voltage is prevented from developing on the transistors.

Description:
BACKGROUND  
         [0001]    1. Field of the Invention  
           [0002]    This invention relates to RF and microwave amplifiers in general and more particularly to Darlington gain block amplifiers that have improved biasing to prevent potentially damaging conditions from occurring. The invention also prevents a voltage pulse on the output of the amplifier that is formed during the start-up process, which can damage the circuitry connected to the output of the gain block amplifier.  
           [0003]    2. Description of Related Art  
           [0004]    Darlington gain block amplifiers are widely used in RF and microwave systems. Darlington amplifiers have a frequency range that starts at DC. The biasing conditions needed by the transistors require de-coupling capacitors at the input and output. The de-coupling capacitors determine the low end of the amplifier frequency range. Normally, Darlington gain block amplifiers receive a DC bias voltage from an ideal current source. In practice, the current source is replaced by a voltage source with a series resistor, which converts it to a reasonably good current source. The resistor value typically ranges up to a few hundred ohms. Therefore, the DC voltage from the voltage source can be up to 10 to 15 volts. An optional choke can be connected in series with the resistor to increase the total impedance at the high end of the amplifier frequency range to minimize gain and power loss.  
           [0005]    Referring to FIG. 1, a typical Darlington gain block amplifier  12  and biasing configuration is shown. A pair of NPN transistors Q 1  and Q 2 , are connected in a Darlington configuration. Transistor Q 1  has a base QB 1 , an emitter QE 1  and a collector QC 1 . Transistor Q 2  has a base QB 2 , an emitter QE 2  and a collector QC 2 . The emitter QE 1  is connected to base QB 2 . The collectors QC 1  and QC 2  are connected together. A biasing network  14  is connected to the amplifier. Biasing network  14  supplies the proper biasing voltages to transistors Q 1  and Q 2 . Resistor R 1  is connected between base QB 1  and collector QC 1 . Resistor R 2  is connected between base QB 1  and ground. Resistor R 3  is connected between emitter QE 1  and ground. Resistor R 4  is connected between emitter QE 2  and ground. A 50 ohm resistor R 5  and input de-coupling capacitor C 1  are serially connected between base QB 1  and ground. 50 ohm resistor R 6  and output de-coupling capacitor C 2  are serially connected between the collectors QC 1 , QC 2  and ground. A controlled current source P 1  is connected to the collector QC 1 , QC 2  junction. The input to the amplifier is on the base QB 1 . The output from the amplifier is taken from the collector QC 2 .  
           [0006]    If a low frequency response is desired, the de-coupling capacitors C 1  and C 2  can have a significant value. For example, for a frequency of 100 KHz, the de-coupling capacitors would have a value around 0.2 μF or more. Where the input and output impedances are equal to a value of 50 ohms, the input and output de-coupling capacitors normally have the same value.  
           [0007]    Unfortunately, an initial start-up process can result in a dangerous ‘voltage bump’ or overshoot voltage occurring on the transistors of the Darlington amplifier. In addition to the danger of damaging the Darlington amplifier, the voltage bump can overload and possibly damage the circuitry connected to the output of the Darlington amplifier. The transient process when the current source has a sharp ramp (small rise time) will be detailed next. The worst case is when the current source P 1  goes from zero to the nominal current value instantaneously. This would occur if the current source was manually connected such as by plugging in a connector or by turning on a mechanical switch. Initially, the de-coupling capacitors C 1  and C 2  are not charged and transistors Q 1  and Q 2  are not conducting.  
           [0008]    The initial response of the circuit of FIG. 1 can be analyzed by using an example of a simplified circuit. Turning to FIG. 2, a simplified circuit  20  of FIG. 1 is shown. All the component values in FIG. 2 are typical values for a Darlington amplifier. FIG. 2 has two current probes  11  and  12 , current source P 1 , 510 ohm resistor R 1 , 580 ohm resistor R 2 , 50 ohm resistor R 5 , 50 ohm resistor R 6  and 0.2 μF capacitors C 1  and C 2 . Current source P 1  has a rise time of 100 nanoseconds. The circuit of FIG. 2 was analyzed using Agilent ADS 2001 circuit simulator software. When the current source is turned on, capacitors C 1  and C 2  will begin to charge. However, capacitors C 1  and C 2  will charge at a differing rate due to the difference in time constants because series resistors R 1 +R 5  and R 6  are different. The total current will be split into two unequal parts, initially in a proportion to approximately (R1+R5)/R6.  
           [0009]    [0009]FIGS. 3 and 4 show the voltage versus time for the simplified circuit  20  of FIG. 2 at nodes ‘DC out’ and ‘base’, respectively. FIG. 5 shows current versus time for circuit  20  at probes I 1  and I 2 . As seen in FIG. 5, the current I 1  charging C1 is about 10 times less than the current charging C2. During charging, the current source split ratio will change because a larger portion of the current will sink through R1 and R2 directly to ground. After more than 1 millisecond, C2 will be charged and its charge current I 2  will drop to almost zero in FIG. 5. At this point, input capacitor C 1  will also be charged and all of the 17 milliamps of current will sink through resistors R 1  and R 2 . The total voltage drop across resistors R 1  and R 2  is 18.5 volts as seen in FIG. 3 and this is equivalent to the maximum voltage on node ‘DC out’. The voltage drop across resistor R 2  is 9.9 volts as seen in FIG. 4. This voltage is equivalent to the maximum voltage on the node ‘base’ of transistor Q 1  of circuit  20 .  
           [0010]    When transistors Q 1  and Q 2  are added to the simulated circuit, they will start to conduct when the voltage across R 2  reaches 2.6 to 2.8 volts. This is twice the base to emitter voltage drop (Vbe) for an indium phosphide heterojunction bi-polar transistor (InP HBT). Transistor Q 1  starts to conduct when the node base voltage reaches approximately 1.3 to 1.4 volts. At this point, the current sinking through transistor Q 1  will not be high. Transistor Q 2  starts to conduct when the node base voltage reaches approximately 2.6 to 2.8 volts and sinks most of the current. At the time that transistor Q 2  starts to conduct, the device voltage will already be 7 volts. Therefore, when transistor Q 2  first starts to conduct it will be subjected to the 7 volts peak voltage. Note that the total peak current through the amplifier of FIG. 8 exceeds the nominal current source value of 17 ma because the discharging current of output capacitor C 2 . If the transistors survive the overvoltage and discharge current conditions, they will shunt the current until the device voltage drops to a steady value of 5 volts. This is shown in FIGS.  6 - 9 . FIG. 6 shows the total device voltage at node ‘DC out’ versus time.  
           [0011]    [0011]FIG. 7 shows the amplifier input voltage or the base voltage of input transistor Q 1  versus time.  
           [0012]    [0012]FIG. 8 shows the total device current versus time, which includes the current from the current source and the discharge current from output capacitor C 2 . This can be more than the total current source current.  
           [0013]    [0013]FIG. 9 shows the emitter currents through transistors Q 1  and Q 2  versus time.  
           [0014]    The most dangerous moment for the transistors is when the total device voltage reaches a maximum value (marker M 5  in FIG. 6) and transistor Q 2  starts to conduct (marker M 4  in FIG. 9). The total device voltage is about 6.85 volts at this moment.  
           [0015]    FIGS.  10 - 13  show actual oscilloscope measurements on a circuit of FIG. 1 that was built. A Keithley 236 power supply was used for the current source. An Infiniium HP 54035A oscilloscope from Agilent was used to measure the voltage versus time for various current source values. FIG. 10 shows the output voltage versus time for a current source of 11 milli-amps. FIG. 11 shows the output voltage versus time for a current source of 12 milli-amps. FIG. 12 shows the output voltage versus time for a current source of 13 milli-amps. FIG. 13 shows the output voltage versus time for a current source of 14 milli-amps.  
           [0016]    FIGS.  10 - 13  show how quickly the amplifier peak voltage rises with an increase in the current source value. As seen in FIG. 13, permanent damage to the output transistors of the amplifier occurred with a current source value of 14 milli-amps. The amplifier device voltage does not come to a steady 5 volt value. Sequential gain measurements confirm that the amplifier is permanently damaged.  
           [0017]    This problem only exists for a current source with small rise times. For a slow current source, the charging rate of the input capacitor C 1  and output capacitor C 2  are about the same and determined mostly by the long rise time of the current source. Therefore the output voltage will not have the peaking shape and the voltage overshoot or voltage bump problem does not occur. With a slow current source, the amplifier can typically handle up to 51 milli-amps of current. Therefore, the overshoot problem is a pure transient issue at start up. It is not a steady state power dissipation limit problem.  
           [0018]    While various Darlington transistor biasing schemes have previously been used, they have suffered from not being able to adequately protect against destructive transient voltage overshoot conditions and from interfering with normal amplifier operation. A current unmet need exists for an improved Darlington amplifier biasing circuit that is low in cost, protects against destructive transient voltage overshoot conditions and does not interfere with normal amplifier operation.  
         SUMMARY  
         [0019]    It is a feature of the invention to provide a Darlington amplifier circuit that prevents destructive transient voltage overshoot conditions from occurring.  
           [0020]    Another feature of the invention is to provide an amplifier circuit that includes a first transistor having a first base, a first collector and a first emitter. A second transistor has a second base, a second collector and a second emitter. The emitter of the first transistor is connected to the base of the second transistor. The collector of the first transistor is connected to the collector of the second transistor. A first resistor is connected between the collector of the first transistor and the base of the first transistor. A second resistor is connected between the base of the first transistor and ground. A first capacitor is connected to the base of the first transistor. A second capacitor is connected to the collectors of the first and second transistors. The second capacitor has a capacitance value at least 5 times larger than the first capacitor. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0021]    [0021]FIG. 1 is a schematic diagram of a prior art Darlington gain block amplifier.  
         [0022]    [0022]FIG. 2 is a schematic of a simplified circuit of FIG. 1.  
         [0023]    [0023]FIG. 3 is a graph of voltage at node ‘dc out’ versus time for the simplified circuit  20  of FIG. 2.  
         [0024]    [0024]FIG. 4 is a graph of voltage at node ‘base’ versus time for the simplified circuit  20  of FIG. 2.  
         [0025]    [0025]FIG. 5 is a graph of current versus time for the simplified circuit  20  of FIG. 2 showing charging of capacitors C 1  and C 2 .  
         [0026]    [0026]FIG. 6 is a graph of the total amplifier device voltage versus time for circuit  12  of FIG. 1.  
         [0027]    [0027]FIG. 7 is a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  12  of FIG. 1.  
         [0028]    [0028]FIG. 8 is a graph of the total device current versus time for circuit  12  of FIG. 1.  
         [0029]    [0029]FIG. 9 is a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  12  of FIG. 1.  
         [0030]    [0030]FIG. 10 is a graph of the oscilloscope measurement of the amplifier output voltage versus time for a current source of 11 milli-amps for circuit  12  of FIG. 1.  
         [0031]    [0031]FIG. 11 is a graph of the oscilloscope measurement of the amplifier output voltage versus time for a current source of 12 milli-amps for the circuit  12  of FIG. 1.  
         [0032]    [0032]FIG. 12 is a graph of the oscilloscope measurement of the amplifier output voltage versus time for a current source of 13 milli-amps for the circuit  12  of FIG. 1.  
         [0033]    [0033]FIG. 13 is a graph of the oscilloscope measurement of the amplifier output voltage versus time for a current source of 14 milli-amps for the circuit  12  of FIG. 1.  
         [0034]    [0034]FIG. 14 is a graph of the total amplifier device voltage versus time for circuit  12  of FIG. 1 and the value of C2 is 5 times larger than C1.  
         [0035]    [0035]FIG. 15 is a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  12  of FIG. 1 and the value of C2 is 5 times larger than C1.  
         [0036]    [0036]FIG. 16 is a graph of the total device current versus time for circuit  12  of FIG. 1 and the value of C2 is 5 times larger than C1.  
         [0037]    [0037]FIG. 17 is a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  12  of FIG. 1 and the value of C2 is 5 times larger than C1.  
         [0038]    [0038]FIG. 18 is a graph of the total amplifier device voltage versus time for circuit  12  of FIG. 1 and the value of C2 is 10 times larger than C1.  
         [0039]    [0039]FIG. 19 is a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  12  of FIG. 1 and the value of C2 is 10 times larger than C1.  
         [0040]    [0040]FIG. 20 is a graph of the total device current versus time for circuit  12  of FIG. 1 and the value of C2 is 10 times larger than C1.  
         [0041]    [0041]FIG. 21 is a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  12  of FIG. 1 and the value of C2 is 10 times larger than C1.  
         [0042]    [0042]FIG. 22 is a graph of the total amplifier device voltage versus time for circuit  12  of FIG. 1 and the value of C2 is 484 times larger than C1.  
         [0043]    [0043]FIG. 23 is a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  12  of FIG. 1 and the value of C2 is 484 times larger than C1.  
         [0044]    [0044]FIG. 24 is a graph of the total device current versus time for circuit  12  of FIG. 1 and the value of C2 is 484 times larger than C1.  
         [0045]    [0045]FIG. 25 is a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  12  of FIG. 1 and the value of C2 is 484 times larger than C1.  
         [0046]    [0046]FIG. 26 is a graph of the actual oscilloscope measurement of the amplifier output voltage versus time for circuit  12  of FIG. 1 that was built with C1=0.013 uF and C2=6.3 uF (C2 is 484 times larger than C1).  
         [0047]    [0047]FIG. 27 is a schematic diagram of an alternative embodiment of the present invention.  
         [0048]    [0048]FIG. 28 is a graph of the total amplifier device voltage versus time for circuit  200  of FIG. 27.  
         [0049]    [0049]FIG. 29 is a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  200  of FIG. 27.  
         [0050]    [0050]FIG. 30 is a graph of the total device current versus time for circuit  200  of FIG. 27.  
         [0051]    [0051]FIG. 31 is a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  200  of FIG. 27.  
         [0052]    [0052]FIG. 32 is a graph of the currents IR 1  through resistor R 1  and currents ID 1 , ID 2  through diodes D 1  and D 2  versus time for circuit  200  of FIG. 27.  
         [0053]    [0053]FIG. 33 is a schematic diagram of another embodiment of the present invention.  
         [0054]    [0054]FIG. 34 is a graph of the total amplifier device voltage versus time for circuit  300  of FIG. 33.  
         [0055]    [0055]FIG. 35 is a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  300  of FIG. 33.  
         [0056]    [0056]FIG. 36 is a graph of the total device current versus time for circuit  300  of FIG. 33.  
         [0057]    [0057]FIG. 37 is a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  300  of FIG. 33.  
         [0058]    [0058]FIG. 38 is a graph of the current IR 1  through resistor R 1  and current IQ 3  through transistor Q 3  versus time for circuit  300  of FIG. 33.  
         [0059]    [0059]FIG. 39 is a schematic diagram of another embodiment of the present invention.  
         [0060]    It is noted that the drawings of the invention are not to scale. In the drawings, like numbering represents like elements between the drawings.  
     
    
     DETAILED DESCRIPTION  
       [0061]    One possible way to prevent the voltage overshoot problem on transistors Q 1  and Q 2  is to use a slow power supply at start-up. This solution to use a special power supply is not practical for most applications. In many cases, slow start up is not acceptable from the overall system requirements and restrictions. It is also expensive to design this type of power supply.  
         [0062]    The voltage overshoot on transistors Q 1  and Q 2  at start up can be controlled by varying the capacitance ratio of capacitor C 2  to C 1 . The ratio C2/C1 should be equal or greater than (R1+R5)/R6. This will equalize the time constant of C1*(R1+R5) and C2*R6 and therefore equalize the charging rate of capacitors C 1  and C 2 . For the given example, If the value of C2 is made 5 to 10 times larger than the value of C1, the voltage overshoot on transistors Q 1  and Q 2  can be prevented.  
         [0063]    Referring to FIGS.  14 - 17 , a simulation is shown for circuit  12  of FIG. 1 with the value of capacitor C 2  five times larger than C1. FIG. 14 is a graph of the total amplifier device voltage versus time. The ‘voltage bump’ or overshoot in FIG. 14 is much less than that in FIG. 6 where the value of capacitors C 1  and C 2  are equal. FIG. 15 is a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  12  of FIG. 1. FIG. 16 is a graph of the total device current versus time for circuit  12  of FIG. 1. There is no ‘current bump’ or current overshoot in FIG. 16 compared with the significant ‘current bump’ shown in FIG. 8 where the values of capacitors C 1  and C 2  are equal. The graph of FIG. 16 shows that capacitor C 2  does not accumulate an extra charge, which can discharge through transistor Q 2  as it turns on causing damage.  
         [0064]    [0064]FIG. 17 shows a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  12  of FIG. 1. As can be seen in FIG. 17, the current IQ 2  through transistor Q 2  does not have the ‘current bump’ because capacitor C 2  does not accumulate extra charge.  
         [0065]    Referring to FIGS.  18 - 21 , a simulation is shown for circuit  12  of FIG. 1 with the value of capacitor C 2  ten times larger than C1. The ratio of C2/C1 is approximately equal to (R1+R5)/R6. FIG. 18 is a graph of the total amplifier device voltage versus time. The ‘voltage bump’ or overshoot voltage effect is further reduced. FIG. 19 is a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  12  of FIG. 1. FIG. 20 is a graph of the total device current versus time for circuit  12  of FIG. 1. The ‘current bump’ has been eliminated. FIG. 21 is a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  12  of FIG. 1. As can be seen in the graphs, the ‘current bump’ through transistor Q 2  and therefore through the amplifier is eliminated. The ‘voltage bump’ or overshoot is also eliminated.  
         [0066]    Referring to FIGS.  22 - 25 , a simulation is shown for circuit  12  of FIG. 1 with the value of capacitor C 2  484 times larger than C1. In this case, the ratio of capacitor C 2  to C 1  is further increased and the ratio is much more than (R1+R5)/R6. In this example C1=0.013 uF and C2=6.3 uF. FIG. 22 is a graph of the total amplifier device voltage versus time for circuit  12  of FIG. 1. There is no ‘voltage bump’ or overshoot present. FIG. 23 is a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  12  of FIG. 1. FIG. 24 is a graph of the total device current versus time for circuit  12  of FIG. 1. There is no ‘current bump’ or overshoot present. FIG. 25 is a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  12  of FIG. 1. As can be seen in FIGS.  22 - 25 , increasing the ratio of capacitor C 2  to C 1 , beyond the ratio of (R1+R5)/R6 slightly reduces the already insignificant voltage overshoot that was shown in FIGS.  18 - 21 .  
         [0067]    [0067]FIG. 26 shows an actual oscilloscope measurement for the circuit  12  of FIG. 1 that was built with C1=0.013 uF and C2=6.3 uF. FIG. 26 shows that the overshoot problem has been eliminated.  
         [0068]    Varying the ratio of the input de-coupling capacitor to the output decoupling capacitor can be done for various Darlington amplifier configurations other than those shown in FIG. 1. This technique can also be applied to non Darlington amplifier configurations including but not limited to pulse amplifiers and video amplifiers.  
         [0069]    Turning now to FIG. 27, another embodiment of the present invention is shown. Darlington amplifier circuit  200  has been modified from that shown in FIG. 1. In circuit  200 , a pair, for example, of serially connected diodes D 1  and D 2  have been connected across resistor R 1 . There can be a single diode, a pair or more diodes connected in series. The number of diodes determines the level of protection of the amplifier. Diodes D 1  and D 2  are shown as transistors connected as diodes. The emitter of D1 is connected to the base QB 1  and resistor R 1  junction. The collector of D2 is connected to the collector QC 1 , resistor R 1  junction. The total voltage drop across the diodes D 1  and D 2  is slightly higher than the voltage drop across resistor R 1  in a steady state. At the beginning of the voltage overshoot, the voltage drop across resistor R 1  increases above the steady state voltage drop on it and will exceed the opening voltage of the diode pair D 1  and D 2 . At this point, diodes D 1  and D 2  start to conduct a current and provide some additional current causing capacitor C 1  to charge faster. This prevents the overshoot voltage. In the steady state, diodes D 1  and D 2  do not conduct and are not interfering with the normal RF operation of the amplifier.  
         [0070]    FIGS.  28 - 32  show simulated results for circuit  200  in FIG. 27. FIG. 28 is a graph of the total amplifier device voltage versus time for circuit  200  of FIG. 27. The ‘voltage bump’ is significantly less than in FIG. 6. FIG. 29 shows a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  200  of FIG. 27. FIG. 30 is a graph of the total device current versus time for circuit  200  of FIG. 27. The current bump or overshoot current is significantly less than in FIG. 8. FIG. 31 is a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  200  of FIG. 27. The ‘current bump’ through transistor Q 2  is significantly less than that shown in FIG. 9. FIG. 32 is a graph of the current IR 1  through resistor R 1  and ID 1 , ID 2  through diodes D 1  and D 2  versus time for circuit  200  of FIG. 27. The peak current ID 1 , ID 2  is almost same value as the peak current IR 1  through the resistor R 1 . Therefore, the total peak current charging capacitor C 1  is almost double the value shown in the circuit of FIG. 1. As can be seen in FIGS.  28 - 31 , the overshoot problem is significantly reduced by circuit  200  as compared to circuit  12 .  
         [0071]    Turning now to FIG. 33, another embodiment of the present invention is shown. Darlington amplifier circuit  300  of FIG. 33 has been modified from that shown in FIG. 1. In circuit  300 , a third transistor Q 3  has been connected across resistor R 1 . Transistor Q 3  has a base QB 3 , an emitter QE 3  and a collector QC 3 . The emitter of transistor Q 3  is connected to the junction of base QB 1  and resistor R 1 . The collector of transistor Q 3  is connected to the junction of collector QC 1  and resistor R 1 . The base of transistor Q 3  is connected to a voltage divider consisting of resistors R 7  and R 8 . Resistor R 7  is connected to node the junction of collector QC 1 , QC 3  and resistor R 1 . Resistor R 8  is connected to ground. The ratio of resistors R 7  and R 8  is chosen based on the maximum required RF peak voltage on the output of the amplifier. The RF peak voltage at the 1 dB compression point determines the resistor R 7 , R 8  division ratio.  
         [0072]    The resistor R 7 , R 8  division ratio should be set such that the 1 dB compression point degrades by only 0.1 to 0.2 dB at low frequencies with reference to the 1 dB compression point of the same amplifier but without the protective circuit.  
         [0073]    In the steady state, transistor Q 3  is not conducting because the voltage on the base of transistor Q 3  set by the voltage divider of R7 and R8 is less than the opening voltage Vbe for transistor Q 3 . Therefore, transistor Q 3  does not interfere with the normal RF operation of the amplifier.  
         [0074]    During the start up process, the voltage on the base of transistor Q 3  increases, turning on transistor Q 3 , which causes capacitor C 1  to charge faster. This prevents the overshoot voltage problem. The circuit  300  of FIG. 33 has the added convenience and ability to finely adjust the voltage protective level by varying the values of resistors R 7  and R 8 . As a side effect, the 1 dB compression point of the amplifier may be slightly degraded by about 0.2 dB less.  
         [0075]    FIGS.  34 - 38  show simulated results for circuit  300 . FIG. 34 shows a graph of total amplifier device voltage versus time for circuit  300  of FIG. 33. The ‘voltage bump’ is further reduced as compared to that shown in FIG. 28. FIG. 35 is a graph of the amplifier input voltage or the base voltage of transistor Q 1  versus time for circuit  300  of FIG. 33. FIG. 36 is a graph of the total device current versus time for circuit  300 . The ‘current bump’ is further reduced from that shown in FIG. 30.  
         [0076]    [0076]FIG. 37 shows a graph of the emitter currents IQ 1  and IQ 2  through transistors Q 1  and Q 2  versus time for circuit  300  of FIG. 33. The ‘current bump’ through transistor Q 2  as well as the ‘total bump’ of the total amplifier current is significantly less than in FIG. 31. FIG. 38 is a graph of the current IR 1  through resistor R 1  and current IQ 3  through transistor Q 3  versus time for circuit  300 . The peak current IQ 3  is almost the same value as the peak current IR 1  through resistor R 1 . Therefore, the total peak current charging capacitor C 1  is almost double that shown in circuit  12  of FIG. 1.  
         [0077]    As can be seen in FIGS.  34 - 37 , the overshoot problem is significantly reduced by circuit  300 .  
         [0078]    Referring to FIG. 39, another embodiment of the present invention is shown. Darlington amplifier circuit  400  has been modified from that shown in FIG. 1. In circuit  400 , transistor Q 4  has been connected in parallel across transistor Q 1 . Transistor Q 4  has a base QB 4 , an emitter QE 4  and a collector QC 4 . The emitter QE 4  of transistor Q 4  is connected to the emitter QE 1  of Q1. The collector QC 4  of transistor Q 4  is connected to the collector QC 1  of Q 1 . The base QB 4  of transistor Q 4  is connected to a voltage divider consisting of resistors R 7  and R 8 . Resistor R 7  is connected to the collectors of Q 1  and Q 4 . Resistor R 8  is connected to ground.  
         [0079]    The ratio of resistors R 7  and R 8  is chosen based on the maximum required RF peak voltage on the output of the amplifier. The RF peak voltage at the 1 dB compression point determines the resistor R 7 , R 8  division ratio.  
         [0080]    The resistor R 7 , R 8  division ratio should be set such that the 1 dB compression point degrades by only 0.1 to 0.2 dB at low frequencies with reference to the 1 dB compression point of the same amplifier but without the protective circuit.  
         [0081]    During steady state operation, transistor Q 4  does not interfere with the normal RF operation of the amplifier because the base to emitter voltage of transistor Q 4 , which is set by the resistor divider R 7 , R 8  is below the voltage required to open transistor Q 4 .  
         [0082]    It is noted that circuit  200  would have the least influence on the RF performance of the Darlington amplifier. The circuit changes shown in circuits  200  and  300  can be readily implemented on existing Darlington amplifiers by adding external circuitry. These circuit changes can also be implemented on a semiconductor die during fabrication.  
         [0083]    The present invention has several advantages. Changing the ratio of the input to output de-coupling capacitors provides a solution to the problem of voltage overshoot on Darlington amplifiers without extra space needs and the expense of additional components. The three circuit modifications that were shown can also be readily implemented at low cost and uses a minimum amount of additional circuit board space to prevent voltage overshoot on Darlington amplifiers. The invention provides an increase in Darlington amplifier reliability and durability at low cost.  
         [0084]    While the invention has been taught with specific reference to these embodiments, someone skilled in the art will recognize that changes can be made in form and detail without departing from the spirit and the scope of the invention. The described embodiments are to be considered in all respects only as illustrative and not restrictive. The scope of the invention is, therefore, indicated by the appended claims rather than by the description. All changes that come within the meaning and range of equivalency of the claims are to be embraced within their scope.