Abstract:
Direct digital QAM modulation at an RF frequency is obtained from digitally synthesized RF signals which are generated for use as the two vectors. The two vectors are individually controlled in phase and summed to provide a combined phase and amplitude modulation that forms the modulated signal. The synthesized RF signal is generated from a higher reference frequency using a variable pulse stretching technique implemented with programmable delay lines. The amount of the pulse stretch in each cycle is controlled by a phase increment value. Pulse stretching can be extended beyond one cycle by pulse swallowing, allowing the generation of an RF signal from DC up to and including the input reference frequency. Phase modulation is added by digital control of the pulse stretching according to the phase modulation data bits.

Description:
This application claims priority under 35 U.S.C. 119 from Provisional Application Ser. No. 60/513,985 filed Oct. 27, 2003. 
   This invention relates generally to telecommunication systems. The present invention relates more specifically to a method of synthesizing a direct QAM modulated RF signal with high power efficiency for use in telecommunication systems. 
   RELATED APPLICATIONS 
   This application is related to applications filed on the same day by the same inventors under, Ser. No. 10/796,415, entitled APPARATUS FOR FRACTIONAL RF SIGNAL SYNTHESIS WITH PHASE MODULATION and, Ser. No. 10/796,417, entitled METHOD AND APPARATUS FOR FRACTIONAL RF SIGNAL SYNTHESIS the disclosures of which are incorporated herein by reference. 

   BACKGROUND OF THE INVENTION 
   A prior art arrangement is shown and described hereinafter and has a number of disadvantages which will become apparent from the description hereinafter. 
   A search has revealed the following US patent references: 
   U.S. Pat. No. 5,329,259 Stengel, “Efficient Amplitude/Phase Modulation Amplifier” 
   U.S. Pat. No. 5,612,651 Chethik, “Modulating Array QAM Transmitter” 
   U.S. Pat. No. 5,659,272 Linguet, “Amplitude Modulation Method and Apparatus using Two Phase Modulated Signals” 
   U.S. Pat. No. 5,867,071 Chethik, “High Power Transmitter Employing a high Power QAM Modulator” 
   U.S. Pat. No. 6,366,177 McCune, “High-Efficiency Power Modulators” 
   U.S. Pat. No. 5,852,389 Kumar, “Direct QAM Modulator” 
   SUMMARY OF THE INVENTION 
   According to the invention there is provided an apparatus for directly generating a QAM RF signal comprising: 
   a high speed reference clock providing in an input signal having a series of pulses at a frequency of the reference clock which is higher than the desired output frequency; 
   two programmable digital delay elements each arranged to receive the reference pulses of the input reference clock and to generate therefrom using input data a respective one of two digital vectors; 
   and a signal combining element for receiving the digital vectors from the programmable digital delay elements and for generating the QAM RF signal therefrom. 
   Preferably there are provided amplifiers for amplifying the digital vectors non linearly before combining. 
   Preferably the programmable digital delay elements comprise high speed adders/accumulators wherein said adders/accumulators are arranged to determine the amount of delay implemented by the delay elements on the reference signal. 
   Preferably the output frequency is set from an increment value according to the following equation:
 
Increment Value=(( f   ref   /f   out )−1)*2 n  
 
where f ref =Reference clock ( 103 ) frequency
 
   f out =Output ( 110 ) frequency 
   n=Number of bits in the accumulator math. 
   Preferably the duty cycle is set by initializing the difference of the initializing values of the two accumulators according to the following equation: 
   The reference clock frequency divided by the desired output frequency multiplied by 2 n  multiplied by (p/100), where p is the percentage duty cycle and n is the number of bits in the accumulator math. 
   Preferably the worst case frequency resolution is determined by the equation: 
   The reference frequency divided by 2 n  where n is equal to the number of bits in the accumulator. 
   Preferably a non-linear amplifier is used to produce a high RF output power, from the sum of two phase modulated vectors. 
   Preferably the duty cycle of the output can be varied by changing the difference in the start values of the accumulators for the rising and falling edge delay control. 
   Preferably the interpolator is a linear interpolator. 
   Preferably the interpolator is a (sin x)/x interpolator filter. 
   Preferably the need for a reconstruction filter is removed by interpolation up to the reference clock rate. 
   Preferably phase delay of the programmable delay is calibrated using a look up table or Microprocessor. 
   Preferably separate delay controls are used for producing the rising and falling edges of the output from the same input edge of the reference clock. 
   Preferably the reference edge of the reference clock is delayed by the programmable delay lines. 
   Preferably the reference edge may be either the rising or falling edge of the reference clock. 
   Preferably the carry bits (overflow bits) are used to control a pulse swallowing circuit to extend the delay to multi cycles of the input reference clock. 
   Preferably the clock swallow circuit can ignore/block multiple reference clock pulses thus giving the delay line endless delay capability. 
   Preferably the clock swallow circuit can be located prior to or following the programmable delay line. 
   Preferably a set reset flipflop is used to combine the separate rising and falling edge delays to form any desired duty cycle output. 
   Preferably the output duty cycle is not dependent on the input duty cycle. 
   Preferably increasing the number of bits in the adder math increases the frequency resolution with negligible degradation in the phase noise performance. 
   Preferably the number of bits of math used in the adder can be equal to or exceed the number of bits of control in lookup table and/or the programmable delay. 
   Preferably the speed can be increased using parallel processing in the adders, and/or accumulators. 
   Preferably the adders/accumulators can be implemented in a larger lookup table wherein all the answers of the pattern are pre-computed and stored. 
   Preferably an optional arrangement could include plurality of adders, accumulators, pulse swallow circuits, lookup tables, and programmable delay lines are used. 
   Preferably the lookup table has a multiple set of lookup tables to be used for temperature compensation of the programmable delay line. 
   Preferably the implementation is done fully digitally in an ASIC with no requirement for a voltage controlled oscillator, loop filter, or Digital to Analog converter used in prior art solutions. 
   Preferably an optional arrangement could include filtering of the output to produce a signal having less harmonics. 
   Thus the arrangement described herein pertains to a new method and apparatus to produce a fully digital QAM modulated frequency agile RF signal. It is based on the summation of two fixed amplitude digital vectors each of which is synthesized from a high fixed-frequency reference clock. Pulse stretching is used to delay each edge of the reference clock to the desired time. Clock edges are swallowed in conjunction with the delay to reproduce the clock edge that synthesizes any desired lower frequency. Phase modulation of the two signal vectors is achieved through the control of the delay with the modulating signal. The invention results in direct high output power, high frequency resolution, low phase noise, wide frequency setting ability, and fully digital ASIC implementability. It also results in superior power efficiency performance. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a block diagram of a Prior Art IQ QAM modulator. 
       FIG. 2  is a block diagram of a System for QAM RF signal synthesis according to the present invention. 
       FIG. 3  is a Timing diagram for the Sample shown in Table 1. 
       FIG. 4  is a graph showing a Sampled Baseband Spectrum in the system of  FIG. 2 . 
       FIG. 5  is a graph showing a Linearly Interpolated Baseband Spectrum the system of  FIG. 2 . 
       FIG. 6  is a graph showing a Sampled Baseband Frequency Spectrum the system of  FIG. 2 . 
   

   Table 1 shows Sample timing calculations for single Vector of the present Invention. 
   DETAILED DESCRIPTION 
   A prior art, architecture  15  for a QAM modulator  17  is shown in  FIG. 1 . The modulator  17  accepts a digital input  19  which is fed to an encoder  23 . The encoder  23  divides the incoming signal into a symbol constellation corresponding to in-phase (I) (x r (nT)) and quadrature (Q) (jx i (nT)) phase components while also performing forward error correction (FEC) required for subsequent decoding in the demodulator. The converter&#39;s outputs are fed to the two identical finite impulse response (FIR) square-root raised Nyquist matched filters  25 ,  27 . These Nyquist filters  25 ,  27  are a pair of identical interpolating low-pass filters which receive the I (X r (nT)) and Q (jx i (nT)) signals from the encoder  23  and generate real and imaginary parts of the complex band-limited baseband signal. The Nyquist filters  25 ,  27  ameliorate intersymbol interference (ISI) which is a by-product of the amplitude modulation with constrained bandwidth. After filtering, the in-phase (y r (nT′)) and quadrature (y i (nT′)) components are multiplied  29 ,  31  with the IF centered carrier signals  33 ,  35 . The multiplied signals are then summed  37  producing a band limited IF QAM output signal (g(nT)). The digital signal is then converted to an analog signal using a DIA converter  39 . The analogue signal is processed  40  and fed to a linear power amplifier  41  for amplification and transmission at  42 . Due to the limited frequency range of the DIA converter  39 , the analog signal processing  40  may also contain upconversion to convert the IF frequency from the D/A output  39  to an RF frequency signal. This method requires a large, very linear power amplifier as the modulation must be produced at low power. This consequently results in very poor power efficiency. 
   The present device is arranged to synthesize a direct QAM modulated signal digitally. This is achieved by summing two digitally produced phase modulated vectors which together implement the required phase and amplitude modulation for the QAM signal. The amplitude modulation is only generated at the last step so that all previous functions are handled in the digital domain. Therefore, the amplification of each vector can be done by a non linear and very efficient amplifier as each vector has only phase modulation and no amplitude modulation. Further, each modulated vector is produced with high resolution from a fixed-frequency high speed reference clock.  FIG. 2  presents a block diagram of the invention. The high speed reference clock  103  would typically be an external input with high frequency absolute accuracy and very low phase noise performance. Examples of sources are well known in the art and include high frequency crystal oscillators, SAW oscillators, and crystal oscillators with harmonic multiplication. 
   The device delays an edge of the reference clock by an amount which is controlled by the modulation adder  102   a,    102   b  and implemented by the programmable delay  106   a,    106   b.  The reference edge could be either the rising or falling edge of the reference clock. There are separate circuits for the control of the two edges so that the rising and falling edge of the output signal  150  can be independently controlled. This ensures that even if the duty cycle of the input reference is not 50%, the output  150  duty cycle can be controlled as both the rising edge and falling edge delay is triggered from the same edge of the reference clock  103 . The desired output duty cycle is typically 50% to maximize the RF power in the fundamental frequency but any desired duty cycle can be achieved. Duty cycle is controlled by selling the initial value  114   a,    111   b . The frequency of the RF output is selected by loading the increment value  100 . The operation is controlled by two equations. The first equation controls the RF output frequency and it determines the value to be loaded in the increment value register  101 . Given that the high speed adder/accumulator  102   a,    102   b  is comprised of 2 n  bits, where n is the number of bits in the accumulator math, the increment value  101  is given by the following equation:
 
Increment Value=(( f   ref/   f   out )−1)*2 n  
 
   where f ref =Reference clock ( 103 ) frequency 
   f out =Output ( 110 ) frequency 
   n=Number of bits in the accumulator math. 
   Table 1 shows sample calculations for an example where the high speed reference clock  103  is 1000 MHz and the desired output RF frequency is 734.313739 MHz. A value of n=12 with 12 for 12 bit adding operations is used. Using these numbers in the frequency setting equation yields an increment value  101  of 1482. This increment value is added on each clock cycle to the accumulator to produce a new accumulator value. 
   The second equation controls the duty cycle of the output. As shown in  FIG. 2 , there are separate blocks to control the rising edge delay (a) and the falling edge delay (b). To accomplish a fixed duty cycle, the increment values  101  must be the same and the initial start up values  111   a  and  111   b  in the accumulator must be set to provide for the desired fixed delay between them. The equation for the initializing value  111   b  assuming the initializing value for  111   a  to be zero is as follows:
 
Initializing Value (111 b  assuming 111 a  is 0)=( f   ref   /f   out )*2 n *( p/ 100)
 
where f ref =Reference clock  103  frequency
 
   f out =Output  110  frequency 
   n=Number of bits in the accumulator math 
   p=Percentage duty cycle 
   For the example shown in Table 1, for duty cycle p=50%, the initializing value  111   b  is calculated to be 2789. Table 1 illustrates that the adder/accumulator  102   a  starts at 0 and increments 1482 at every rising edge of the clock. At the same time adder/accumulator  102   b  starts at 2789 and increments 1482 every rising edge of the clock. Any phase modulation required is added in a second modulation adder  120 . When the modulation adder  120  overflows and produces a carry out due to the math addition, an input pulse edge must be ignored or “swallowed”. This corresponds to phase wraparound, i.e. the phase shift has reached 360 degrees and must be set to 0 degrees. In the present invention, 2 n  is calibrated to equal 360 degrees of the reference clock input  103 . This calibration is performed in the LUT  105   a,    105   b  by a simple mapping of input control bits to desired control lines. The filling of the LUT  105   a,    105   b  to perform this requirement would be well understood by those skilled in the art. The LUTs  105   a,    105   b  can be implemented using a read only memory or with a microprocessor. The adder/accumulator overflows due to an addition indicates a greater than 360 degree delay requirement. This delay is implemented by using the next clock edge rather than delaying from the original clock edge. This allows the programmable delay line  106   a,    106   b  to act as a delay line with endless delay capability. For example if the accumulator is using 12 bit math then 360 degrees is equal to 2 12  or 4096. In the example shown in Table 1, the accumulator overflows to 4446, which means the overflow bits are set to a value of 1 and accumulator value goes to 4446-4096=350. The circuit implements the requirement for this value of phase delay in two parts. It activates the pulse swallow circuit to ignore one clock edge, and sets the programmable delay to 350 which completes the rest of the delay requirement. This unique feature of the present invention means that any quantity of overflow bits could be handled. lithe addition of the increment value  101  to the accumulator value  102   a,    102   b  causes, for example, two overflow bits, then the pulse swallow circuit  104   a,    104   b  at the output  112  of the accumulator  102   a,    102   b  would ignore or “swallow” 2 pulses. In this way it is possible to synthesis very low frequencies from the high speed clock reference  103 . The delay required to achieve this is limited to one cycle at the high speed reference clock rate. Furthermore, the accuracy of the timing and jitter is excellent, as the time is always relative to the closest edge of the high speed clock reference  103 . The output signal phase noise is not controlled by the loop bandwidth nor the phase noise characteristics of the voltage controlled oscillators applied in traditional methods. Instead, the phase noise performance is directly linked to the high speed reference. This reduces both the jitter and phase noise of the synthesized RF output. The delayed edge from the programmable delay  106   a  sets the output RF high by enabling a set-reset flipflop  107 . When the delayed edge from the programmable delay  106   b  reaches the flipflop, it resets the flip flop  107  and causes the RF output to go low. This completes the synthesis of the RF output at the preferred 50% duty cycle rate.  FIG. 6  illustrates time plots for the example in Table 1. The upper plot is the high speed reference clock plotted over 5500 degrees. The lower plot is the RF output, plotted over that same 5500 degrees of phase shift with respect to the reference clock. The lower plot demonstrated the synthesis of a lower frequency from the high speed reference clock. 
   The frequency step size of this invention depends on the frequency and the number of bits n in the accumulator math. It is coarser at frequencies closer to the reference clock frequency, and finer at lower frequency outputs. The worst case step size is the reference frequency divided by 2 n  where n is equal to the number of bits in the accumulator math. In the example of Table 1, the step size is 1000 MHz divided by 2^ n . This gives a step size of approximately 244 kHz. To improve the frequency resolution an increased number of bits in the math can be used. For example with 16 bit math, the frequency resolution improves to approximately 15.2 kHz. Increasing n to 32 bits would result in approximately 0.2Hz frequency resolution. It is only necessary to increase the number of bits of resolution in the adder/accumulators  102   a,    102   b,  and not necessarily the LUTs  105   a,    105   b  and the programmable dividers  106   a,    106   b.  The remaining least significant bits can be truncated before the LUTs  105   a,    105   b  with negligible effect on the RF output phase noise quality. This means that very fine frequency resolution is achieved with negligible degradation in the phase noise. It can also be seen that the increment values  101  can be changed to provide an essentially instantaneous frequency change. 
   Phase modulation is added by the addition of a second adder  120 . This adder is also high speed and runs at full rate. This modulation adder  120  adds the desired phase offset to the value of the accumulator  102   a ,  102   b  to provide a new increment value that is sent to the look up tables  105   a ,  105   b  and the pulse swallow circuit  104   a ,  104   b.  The number added could be positive or negative. The average value added is always zero over a long period of time. This ensures the overall effect of the modulation adder is only a phase modulation and not a change in the center frequency of operation. Compared to the reference clock frequency, the modulation information ( 122 , 123 ) is at a much lower frequency baseband rate.  FIG. 4  illustrates an example of the incoming sampled baseband using 8 samples per symbol. Graph  200  is the desired phase rate signal control. Graph  201  is the sampled input. If the graph  201  is placed through a reconstruction filter the desired shape  200  will be produced. This is illustrated in spectrum plot of  FIG. 6 . The energy of the sampled waveform  201  is spread over the desired baseband  400  and the clock  404  and aliasing components  402  and  403 . A low pass filter  401  is used in prior art, after a DAC to remove the undesired clock  404  and aliasing components  402  and  403 . However, in the present invention there is no DAC as the phase modulation is achieved by directly adding digitally to the increment value. There is no place to put an analog low pass filter. This would result in clock and aliasing signal components showing up in the RF output  150 . To overcome this problem an interpolator  121  is used to reduce the clock and aliasing signals as well as to shift their frequency so that they may be filtered at the RF output  150  using an optional band bass filter  109 . The preferred embodiment of the interpolator is a linear interpolator. However, it is also valid to use other interpolation techniques such as (sin x)/x interpolation and filtering. Sin x/x interpolation is well understood by those knowledgeable in the art. Linear interpolation is implemented by drawing a straight line between two known points. This is simple to implement as the increment value required for each reference clock cycle is based on the equation: Input sample frequency  122  divided by the clock reference frequency  103  multiplied by the difference of two adjacent sampled data point values. An implementation of the interpolator  121  used for suppressing the clock and aliasing components is shown in  FIG. 5 . The linear interpolated curve  301  now has more power in the desired curve  300  than the non interpolated curve  201 . A full (sin x)/x interpolater would remove the clock and aliasing component as the phase adjust would occur at every reference clock edge. This alleviates the need for any reconstruction filter which is now replaced with a full digital solution that can be implemented using an ASIC. p Another advantage of the present device is that the output signal frequency  150  range is very wide. The pulse swallow  104   a ,  104   b  circuit can block multiple reference clock pulses extending the programmable delay indefinitely. This is only limited by the number of overflow bits and math bits used. The output frequency range coverage can thus be from DC up to the high speed reference clock frequency. It is desirable to have as high a reference clock frequency as possible. A higher reference clock frequency extends the useful frequency range, and improves the frequency resolution. The upper reference frequency limit of the design is mostly limited by the design speeds of the high speed adders/accumulator  102   a ,  102   b  and look up tables  105   a ,  105   b.  It is understood in the art that speeds can be increased by parallel processing and other design techniques. For example multiple high speed adders/accumulator, LUTs or programmable delay lines could be used in parallel for increasing the speed and hence the output signal frequency capability of the invention. 
   The two synthesized RF signals  150  and  154  can be phase modulated independently. The first vector circuit  140  is phase modulated from the bit control inputs of  123  and  122 . The second vector circuit  141  is phase modulated from the bit control inputs of  145  and  146 . These two vector circuits  140  and  141  share the same high speed reference clock  103 , and frequency load increment value ( 100 ). The circuits of  140  and  141  are digital circuits with digital input and outputs. If required, these digital signals can be amplified with  151  and  153  to increase the level of each phase modulated vector. Each vector is still digital and contains no amplitude modulation, so amplification can be done with a non linear, very power efficient amplifiers ( 151  and  153 ), such as a class C amplifier. The output of the amplifiers are combined together in a combiner  152  resulting in an output that has both phase and amplitude modulation. The peak power corresponds to the sum of the two vector powers. The output of the combiner  152  may be optionally filtered  155  to remove harmonics. The result is a phase and amplitude modulated signal  156  that is controlled through the input phase control of Vector A ( 123 ,  122 ) and Vector B ( 145 ,  146 ). The modulation is valid for any level of QAM. 
   Within the spirit of the invention it is also possible to implement the invention on every 180 degrees of the reference clock using both the rising and the falling edges. Another alternative arrangement is to position the clock swallow circuit following the programmable delay line. 
   Within the spirit of the invention it is also possible to remove the adder/accumulators  102   a ,  102   b ) and replace the LUT  105   a ,  105   b  with a larger LUT. A simple counter could increment the values in the LUT. The LUT  105   a ,  105   b  would in this case hold the pre-added values, and just cycle through them until the pattern repeats. 
   Within the spirit of the invention is it also possible to compromise latency for the speed of the device. It does not matter how many clock cycles it takes to implement an adder or LUT for example, as long as we get valid data out every reference clock cycle. 
   It is possible to use a selection of different lookup tables  105   a ,  105   b  or offset values to compensate for the temperature effect on the programmable delay lines  106   a ,  106   b.  It is also possible to vary the implementation of the delay lines by altering the input clock signal. Examples of clock alteration would include frequency multiplication, division, or phase shifting. 
   Since various modifications can be made in my invention as herein above described, and many apparently widely different embodiments of same made within the spirit and scope of the claims without department from such spirit and scope, it is intended that all matter contained in the accompanying specification shall be interpreted as illustrative only and not in a limiting sense.