Abstract:
A transimpedance amplifier includes a current regulator having a first current mirror of a first conduction type and a first current mirror of a second conduction type. The first current mirror stage of a first conduction type receives a reference current. The first current mirror stage of a second conduction type is connected to the first current mirror stage of the first conduction type, and receives an output current from the first current mirror stage of the first conduction type, and to generate a current to be used as a current source by a transimpedance amplifier. Each of the current mirror stages includes a first transistor and a second transistor of the same conduction type having their gate terminals connected wherein the first transistor receives an input current and the second transistor provides an output current that is a factor of the received input current.

Description:
TECHNICAL FIELD 
       [0001]    The present disclosure relates to transimpedance amplifiers that are driven by current sources. It is particularly related to, but in no way limited to, TIAs used for radio frequency (RF) low noise amplifiers (LNA) and analog-to-digital converters (ADC). 
       BACKGROUND 
       [0002]    Transimpedance amplifiers (TIA) are commonly used for providing a voltage output signal proportional to a current signal and are normally implemented by providing a feedback resistor across the input and output nodes of an operational amplifier. TIAs are often known as current to voltage converters. One use of a TIA is to convert an input current signal into an output voltage signal. 
         [0003]      FIG. 1  shows a transimpedance amplifier  100  implemented with MOSFETs. The TIA comprises an n-channel MOSFET (NMOS)  110  and a p-channel MOSFET (PMOS)  120  arranged such that the gate terminals  112 ,  122  of the MOSFETs are tied to an input v in , and the drain terminals  114 ,  124  of the MOSFETs are tied to an output v out . The source terminal  126  of the PMOS is connected to a current source  130  and a discharge capacitor  140 . A feedback resistor  150  is connected across the TIA  100 . With this arrangement, the gain G m  of the amplifier is controlled by the current I S  provided to the TIA by the current source  130  where G m ∝√{square root over (I S )}. 
         [0004]    In RF receiver modules, TIAs generally function as low-noise pre-amplifiers, which largely determine the overall performance of the module. In the past, because of the wide bandwidth and the high gain necessary for sensitive data links, TIAs have been implemented with bipolar and gallium arsenide (GaAs) metal-semiconductor field effect transistors (MESFET). These implementations result in high speed devices, but they can be costly and the production processes lack high-yield manufacturability. Recently, metal-oxide-semiconductor (MOS) technology has become popular for the design of TIAs because of its low cost and high-yield manufacturability. However, using a single MOS gain stage fails to provide enough gain for multi-gigabit operation, because MOS transistors have a lower transconductance than bipolar transistors. Hence, successful high-speed MOS implementations have relied on multiple gain stages. 
         [0005]    At present, the current supplied to a TIA is fixed at a maximum value that ensures the highest gain G m(max)  from the TIA. However, this gain is not always required and so during normal operation of the TIA, this high current is mostly taken up by the load. This unnecessarily increases the power consumption of the TIA when G m(max)  is not required. 
         [0006]    In view of the above, there remains a need to provide a TIA which is driven by a current source that is adaptive to the requirements of the TIA circuit such that a high current source is only provided to a component of the circuit (when G m(max)  is required) while supplying the remaining components of the circuit with lower current sources so as to minimise the overall power consumption. 
       SUMMARY OF THE INVENTION 
       [0007]    This invention relates to a current regulator for generating at least one current source to a current driven transimpedance amplifier. A multi-stage RF transimpedance amplifier comprising a plurality of current driven transimpedance stages is also detailed. 
         [0008]    There is therefore provided a current regulator comprising a first current mirror stage of a first conduction type configured to receive a reference current; and a first current mirror stage of a second conduction type connected to the first current mirror stage of the first conduction type and configured to receive an output current from the first current mirror stage of the first conduction type, and to generate a current to be used as a current source, wherein each of said current mirror stages comprises a first transistor and a second transistor of the same conduction type, said transistors having their gate terminals connected wherein the first transistor is configured to receive an input current and the second transistor is configured to output a current that is a factor of the received input current. 
         [0009]    There is further provided a current regulator wherein the second transistor in the first current mirror stage of the first conduction type comprises a plurality of transistors connected in parallel, said plurality being captured by a replication factor ITRIM that is indicative of the number of transistors connected in parallel that make up the second transistor in the first current mirror stage of the first conduction type. 
         [0010]    Also provided is a current regulator wherein the first transistor in the first current mirror stage of the second conduction type comprises a plurality of transistors connected in parallel, said plurality being captured by a replication factor ISET that is indicative of the number of transistors connected in parallel that make up the first transistor in the first current mirror stage of the second conduction type. 
         [0011]    A current regulator wherein the current generated by the first current mirror stage of the second conduction type is a factor of ITRIM and ISET is also provided. 
         [0012]    Still further, there is provided a current regulator comprising a plurality of further current mirror stages of the second conduction type connected in cascade to the first current mirror stage of the second conduction type wherein each of said plurality of further current mirror stages of the second conduction type generate further currents to be used as further current sources. 
         [0013]    A plurality of further current mirror stages of the first conduction type is also provided where the plurality of further current mirror stages is connected in cascade to the first current mirror stage of the first conduction type wherein each of said plurality of further current mirror stages of the first conduction type feeds a plurality of first current mirror stages of a second conduction type connected in cascade, each of said plurality of first current mirror stages of a second conduction type generating a current to be used as a current source. 
         [0014]    Also provided is a current regulator wherein each of said plurality of further current mirror stages of the first and/or second conduction type has a unique ISET value. Additionally, a current regulator wherein each of said plurality of further current mirror stages of the first conduction type has a unique ITRIM value. Preferably, ISET has a nominal value of 4 and ITRIM has a nominal value of 8, and, optionally, wherein the current regulator generates between 8 to 10 current sources. 
         [0015]    There is further provided a current regulated transimpedance amplifier comprising a first transistor of a first conduction type; a second transistor of a second conduction type opposite to that of the first conduction type, said second transistor coupled to said first transistor; an input terminal for receiving an input signal, wherein the gate terminals of said first and second transistors are coupled to each other and to said input terminal; an output terminal for transmitting an output signal, wherein the output terminal is coupled to said first and second transistors; and a current regulator as described above for supplying a current source to the first transistor. 
         [0016]    Also provided is a multi-stage RF transimpedance amplifier circuit comprising a plurality of transimpedance amplifier stages supplied with current sources generated by a current regulator as described above. 
     
    
     
       FIGURES 
         [0017]      FIG. 1  shows a schematic circuit diagram of a transimpedance amplifier as presently known in the art. 
           [0018]      FIG. 2A  shows a current regulator circuit according to an embodiment of the present invention. 
           [0019]      FIG. 2B  shows a schematic representation of a current regulator circuit used to generate a plurality of current sources according to an embodiment of the present invention. 
           [0020]      FIG. 2C  shows a sensing circuit used by the current regulator circuit according to an embodiment of the present invention. 
           [0021]      FIG. 3  shows a cascade configuration of transimpedance amplifiers driven by a current regulator according to an embodiment of the present invention. 
           [0022]      FIG. 4A  shows a radio frequency receiver circuit comprising the current regulator circuit according to an embodiment of the present invention. 
           [0023]      FIG. 4B  shows an analog-to-digital converter circuit as presently known in the art, comprising the current regulator circuit according to an embodiment of the present invention. 
       
    
    
     DETAILED DESCRIPTION 
       [0024]    An embodiment of the present invention is shown in the current regulator circuit of  FIG. 2A . The current regulator  200  comprises two interconnected current mirrors  210  and  220 . Current mirror  210  comprises NMOS transistors TN 1  and TN 2  and is configured to receive a bandgap reference current i bg . Accordingly, the current flowing through point X in the NMOS current mirror  210  is pi bg , where p is a factor between 22.5 and 27.5. This current then flows through the PMOS current mirror  220 , and results in an output current of I PSRC  as shown in  FIG. 2A . Current I PSRC  is used as a current source PSRC in place of the current source  130  in the TIA shown in  FIG. 1 . 
         [0025]    In one embodiment of the present invention, PMOS transistor TP 1  comprises a plurality of PMOS transistors  250  connected in parallel. The number of PMOS transistors connected in parallel to form transistor TP 1  of the PMOS current mirror  220  gives rise to a parameter ISET. ISET is a replication factor and is indicative of the number of PMOS transistors  250  connected in parallel to form the left-hand branch of the PMOS current mirror  220  as illustrated in  FIG. 2A . Similarly, in a further embodiment of the present invention, the NMOS transistor TN 2  comprises a plurality of NMOS transistors  260  connected in parallel. The number of NMOS transistors connected in parallel to form transistor TN 2  of the NMOS current mirror  210  gives rise to a parameter ITRIM which is a replication factor that is indicative of the number of NMOS transistors  260  connected in parallel to form the right-hand branch of the NMOS current mirror  210  as depicted in  FIG. 2A . 
         [0026]    It can therefore be seen that both ITRIM and ISET enable control of the output current I PSRC , where ISET enables course control of the output current I PSRC  while ITRIM allows fine control (±10%) of output current I PSRC . Accordingly, the magnitude of I PSRC  can be shown to be dependent on ISET and ITRIM according to: 
         [0000]    
       
      
       I 
       PSRC 
       =αβn·i 
       bg  
      
     
         [0000]    where: 
         [0027]    i bg  is the bandgap current (used herein as a reference current), 
         [0028]    n is the number of unit PMOS fingers in parallel in the circuit layout used for the TIA, typically between 6 to 200, 
         [0029]    β is the value set by ISET and is given by β=75/(4×(10−ISET)), and 
         [0030]    α is the value set by ITRIM and is given by α=1+((ITRIM−8)/80). 
         [0031]    In an embodiment of the present invention, the current regulator circuit  200  may be implemented to generate a plurality of current sources. This is achieved by cascading several PMOS current mirror stages similar to PMOS current mirror  220  per NMOS current mirror  210 . An example of such an arrangement is shown in the schematic diagram of  FIG. 2B  which generates current sources PSRC 01  to PSRC 14 . In  FIG. 2B , the current sources are generated in four current source blocks labelled “#1”, “#2”, “#3” and “#4”. Each of the current source blocks has an NMOS current mirror similar to current mirror  210  shown in  FIG. 2A ; NMOS current mirrors  211  to  214  are similar to current mirror  210 ; each of the NMOS current mirrors  211  to  214  have a different number of NMOS transistors connected in parallel that make up transistor TN 2  of  FIG. 2A . Thus, NMOS current mirrors  211  to  214  each have a different associated ITRIM parameter. NMOS current mirrors  211  to  214  are cascaded together where the gate current from one NMOS current mirror is fed into the subsequent NMOS current mirror. In this example, the first NMOS current mirror  211  takes i bg  as the input current. The gate current i G211  of the current mirror  211  is fed into the NMOS current mirror of the next stage  212 , and so on. 
         [0032]    Referring to  FIG. 2B , each NMOS current mirror  211  to  214  feeds at least one PMOS current mirror  221  to  234 ; this has been seen in  FIG. 2A  where NMOS current mirror  210  feeds PMOS current mirror  220 . In the case of the current source blocks of  FIG. 2B , NMOS current mirror  211  of current source block # 1  feeds a cascade of four PMOS current mirrors  221  to  224  where the gate current i G221  from PMOS current mirror  221  is fed into the subsequent PMOS current mirror  222 , for example. This gives rise to current sources PSRC 01  to PSRC 04  in current source block #1. Similarly in current source block #2, NMOS current mirror  212  feeds PMOS current mirrors  225  to  227  connected in cascade, generating current sources PSRC 05  to PSRC 07 . In current source block #3, NMOS current mirror  213  feeds cascaded PMOS current mirrors  228  to  232  giving current sources PSRC 08  to PSRC 12 . In current source block #4, NMOS current mirror  214  feeds cascaded PMOS current mirrors  233  and  234  which generate current sources PSRC 13  to PSRC 14 . 
         [0033]    In each current source block, the PMOS current mirrors  221  to  234  in  FIG. 2B  are similar to PMOS current mirror  220  in  FIG. 2A ; each of the PMOS current mirrors  221  to  234  have a different number of PMOS transistors connected in parallel that make up transistor TP 1  of  FIG. 2A . Thus, the PMOS current mirrors within each current source block each have a different associated ISET parameter, i.e. PMOS current mirrors  221  to  224 ,  225  to  227 ,  228  to  232 , and  233  to  234  each have a different ISET parameter. 
         [0034]    In an exemplified embodiment of the present invention, the current regulator  200  generates fourteen different current sources PSRC 01  to PSRC 14 ; each of these sources has specific functions when used in conjunction with a TIA. This is illustrated in the current source list shown in Table 1. However it will be appreciated that any number of current sources may be generated by the current regulator based on the requirements of the recipient device. 
         [0035]    More current sources may be made available for use by combining the current sources that have been generated. For example, referring to Table 1, PSRC 15  is a derivative current source resulting from the combination of current sources PSRC 01  and PSRC 09 . In Table 1, current sources PSRC 15  to PSRC 20  are derivative current sources created from a combination of the current sources PSRC 01  to PSRC 14  and/or any of derivative current sources PSRC 15  to PSRC 20 . In a further embodiment of the present invention, between 8 to 10 different current sources are generated. In a specific example of the present invention, nominal mid-range values of ISET=4 and ITRIM=8 result in a current of 625 nA per finger of the PMOS transistors used for the TIA. 
         [0036]    Each PMOS current mirror  220  has a sensing circuit  240  for overvoltage and headroom sensing for each of the generated current sources, as shown in  FIG. 2A . The overvoltage sensing protects the voltage regulator circuit  200  against high output voltages from the generated current source. Headroom sensing indicates when there is insufficient headroom (voltage difference) between the power supply and the output voltage from the generated current source. A sensing circuit  240  is shown in  FIG. 2C  and comprises a PMOS transistor  270  connected to a current source  280  and an inverter  290 . Hysteresis of about 10% is applied to the output of the inverter  290 . A reference voltage V REF1  or V REF2  is input to the gate of the transistor  270 . In one embodiment, the current source  280  is implemented as a 90 nA or 100 nA current source. Both types of sensing use the same circuit  240  from the same sense point Q as shown in  FIG. 2A . Headroom sensing is done for each generated current source and uses a VDsat generator to detect a headroom reference voltage V REF1  of less than approximately V REF1 =2V D(sat) . V REF1  is obtained from each PMOS current mirror  220  as shown in  FIG. 2A . When V REF1  is greater than about 2V D(sat) , the respective output of the sensing circuit  240  goes high. Overvoltage sensing is done globally across all current sources using a predefined reference voltage V REF2 . The outputs from the sensing circuit  240  for each of overvoltage and headroom sensing are digital. In the embodiment of the present invention where multiple PMOS current mirrors  220  are used to generate a series of current sources, each having respective sensing circuits  240 , the digital outputs of each of the respective sensing circuits  240  are combined in a NAND manner for each of overvoltage and headroom sensing. 
         [0037]    The current regulator of the present invention is therefore able to generate a series of current sources to be used in TIAs and the like. Such a series of current sources PSRC 01  to PSRC 20  is shown in Table 1, of which PSRC 15  to PSRC 20  are derivative current sources. Exemplary functions of the respective current sources are also listed for illustrative purposes only. 
         [0038]    An implementation of the current regulator of the present invention will now be illustrated with respect to  FIG. 3 . This figure shows a transimpedance amplifier circuit  300  in which multiple TIA stages  310 ,  320  and  330 , each with a gain G m(1) , G m(2)  and G m(3) , are connected in series. Such an arrangement of amplifiers can often be found in analog to digital converters (ADC) and low noise amplifiers (LNA). In  FIG. 3 , each of the TIAs  310 ,  320  and  330  has a corresponding feedback resistor  315 ,  325  and  335 , and a corresponding load resistor  316 ,  326  and  336 . In the implementation shown, the TIAs  310 ,  320  and  330  are driven by respective current sources PSRCx, PSRCy and PSRCz. These current sources are generated by a current regulator  340  such as that described in relation to  FIG. 2A  using PMOS and NMOS current mirrors configured with appropriate ISET and ITRIM values. 
         [0039]    In the configuration shown in  FIG. 3 , the gain G T  of the multi-stage TIA  300  is the product of the individual gains of the cascaded amplifiers, i.e. G T =π i   n G m(i)  for n cascaded stages. Thus in  FIG. 3 , G T  for TIAs  310 ,  320  and  330  is G m(1) ·G m(2) ·G m(3) . It therefore follows that for a multi-stage TIA configuration as shown in  FIG. 3 , a target gain G T  can be attained without requiring the gain G m(i)  of each of the constituent TIAs to be equivalent to G T . In this manner, the current source required to drive the cascaded amplifiers need not be as large as would be required in a single stage TIA configuration; instead the larger current source is only provided by the current regulator to the TIA stage with the higher gain. This minimises on power consumption of the multi-stage TIA. 
         [0040]    From the above, it can be seen that the current regulator of the present invention enables the generation of current sources as required using the appropriate ISET and/or ITRIM replication factor in the respective current mirrors that form the current regulator. Thus when more current is required by a device (such as a TIA), a suitable current source generated by the current regulator is used. This highlights the ability of the current regulator of the present invention to supply suitable current sources on demand. 
         [0041]    A further embodiment of the present invention is shown in  FIG. 4A  which shows the current regulator of the present invention implemented in an RF receiver module  400 . The receiver module  400  comprises LNAs  410  and  420 , and ADCs  430  and  440 . Each of the LNAs  410  and  420  comprise a TIA as previously described. In  FIG. 4A , current regulator  450  is used to supply each of the LNAs and the ADCs with a current source (such as those exemplified in Table 1). This current regulator is similar to that shown in  FIG. 2A . Each LNA  410  and  420  has the configuration as shown in  FIG. 1 . The inner circuitry of each of the ADCs  430  and  440  is shown in  FIG. 4B  and comprises a plurality of cascaded TIAs  460  to  465 . In an embodiment of the present invention, each of the TIAs in the LNAs and the ADCs are supplied by an appropriate current source provided by the current regulator  450 . The current requirements of the LNAs may vary from those of the ADCs; thus the current sources provided by the current regulator  450  to the LNAs are different to those provided to the ADCs. As previously mentioned, the current sources generated by the current regulator  450  may be varied by adjusting the ISET and, if necessary (where more than one current source block is used) ITRIM. These varying requirements of the RF receiver module for current sources of different magnitude are therefore satisfied by a current regulator according to the present invention. By providing various current sources, only a few of which are large in magnitude, the overall power consumption of the RF receiver module is minimised. 
         [0042]    It will be understood that the above description of a preferred embodiment is given by way of example only and that various modifications may be made by those skilled in the art. Although various embodiments have been described above with a certain degree of particularity, or with reference to one or more individual embodiments, those skilled in the art could make numerous alterations to the disclosed embodiments without departing from the spirit or scope of this invention. 
         [0000]    
       
         
               
               
               
             
               
               
               
             
           
               
                 TABLE 1 
               
               
                   
               
               
                 PSRC 
                 Use 
                 μA 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 PSRC01 
                 Ipbuffer 
                 180 
               
               
                 PSRC02 
                 Q Buffer 
                 111 
               
               
                 PSRC03 
                 I Buffer 
                 105 
               
               
                 PSRC04 
                 Spare 
                 0 
               
               
                 PSRC05 
                 Spare 
                 0 
               
               
                 PSRC06 
                 ADC0 and  
                 130 
               
               
                   
                 DAC 
                   
               
               
                 PSRC07 
                 ADC0 bias  
                 10 
               
               
                   
                 cascode 
                   
               
               
                 PSRC08 
                 ADC0  
                 20 
               
               
                   
                 comparator 
                   
               
               
                 PSRC09 
                 Spare 
                 5 
               
               
                 PSRC10 
                 ADC  
                 65 
               
               
                   
                 digital 
                   
               
               
                 PSRC11 
                 ADC1 and  
                 130 
               
               
                   
                 DAC 
                   
               
               
                 PSRC12 
                 ADC1 bias  
                 10 
               
               
                   
                 cascode 
                   
               
               
                 PSRC13 
                 ADC1  
                 20 
               
               
                   
                 comparator 
                   
               
               
                 PSRC14 
                 Spare 
                 0 
               
               
                 PSRC15  
                 Clamp bias 
                 185 
               
               
                 (PSRC01 + PSRC09) 
                   
                   
               
               
                 PSRC16 
                 LNA 0 
                 110 
               
               
                 (PSRC03 + PSRC09) 
                   
                   
               
               
                 PSRC17 
                 LNA 1 
                 120 
               
               
                 (PSRC07 + PSRC16) 
                   
                   
               
               
                 PSRC18  
                 IQCAL and  
                 85 
               
               
                 (PSRC10 + PSRC13) 
                 Rxip 
                   
               
               
                 PSRC19  
                 LNA bias 
                 296 
               
               
                 (PSRC02 + PSRC15) 
                   
                   
               
               
                 PSRC20 
                 Mixbias 
                 195 
               
               
                 (PSRC16 + PSRC18)