Abstract:
A high-frequency resonant sine wave DC to AC inverter suitable for use in a personal computer (PC) power supply includes a full-bridge inverter, a resonant circuit, a phase shift modulation circuit, and a resonant gate driver. The resonant gate driver provides sinusoidal gate drive signals to the full-bridge inverter enabling highly efficient operation on the inverter.

Description:
TECHNICAL FIELD 
     This invention relates to power supplies for electronic equipment and, in particular, to inverters for generating high frequency sinusoidal AC voltages for electronics equipment used in telecommunications and computer systems. Typical examples of potential use are in personal computers, servers, routers, network processors, and opto-electronic equipment. 
     BACKGROUND OF THE INVENTION 
     Segments of the personal computer (PC) industry have dramatically changed during the last decade. The future is even more challenging. A dramatic increase in the processor speeds of PCs has required an overwhelming increase in current and associated dynamics (very high slew rate). This already challenging technical requirement is further complicated by a need for voltage reduction, potentially to sub-volt levels. 
     In the past, there was virtually no challenge in powering computers. A multiple output, very slow power supply called a “Silver Box (SB)” was adapted to meet the requirements of every power demand. However, as silicon development progressed, multiple voltages of less than 3.3V were required. Voltage Regulator Modules (VRMs) on the processor Mother Board (MB) were a logical solution to that problem. Today, the number of VRMs required on the Mother Board is increasing. In addition to the VRMs, a large number of de-coupling capacitors are required in proximity of the processor to meet the requirements of very high slew rate of the current. This has resulted in a rapid increase in the cost, as well as a large reduction in overall efficiency, of the power delivery system. 
     A number of options for improving this situation have been explored. For example, Advanced Voltage Regulator Module (AVRM) offers the capability to supply high di/dt and high current, however, at increased cost, and with low efficiency and moderately high capacity of the de-coupling capacitors. Replacing low voltage DC distribution with higher DC voltage, such as 48V, is more promising but has a drawback of higher cost. Recently a novel High Frequency Alternating Current (HFAC) power delivery architecture has been proposed for powering the future generation PCs in reference entitled, “PC Platform Power Distribution System: Past Application, Today&#39;s Challenge and Future Direction” published in the conference proceedings of International Telecommunications Energy Conference, Copenhagen, Denmark, June 1999 by J. Drobnik, L. Huang, P. Jain and R. Steigerwald. In the HFAC architecture, the system power supply (silver box) generates high frequency and high voltage. The HFAC is then fed to an individual AC-DC converter (ACVRM) and converted into DC of specific parameters at the point of use. 
     HFAC is conceptually the simplest architecture proposed to date, which deals with all of the power delivery issues defined above. This includes elimination of duplicated power conversions, and active energy steering without additional components. 
     The key to successful implementation of an HFAC power delivery system resides in the two stages of power conversion namely; DC to AC high frequency conversion stage and the stage that converts high frequency AC to DC. 
     FIG. 1 shows a block diagram of a conventional DC to high frequency AC inverter  100 . The inverter  100  includes a full-bridge inverter  104  having an input  104 A for receiving a DC input voltage  102  and providing an output  104 B. The output  104 B is connected at  106  to an input  108 A of a resonant circuit  108 . An output  108 B of the resonant circuit  108  provides a high frequency AC output voltage  110 . The AC output voltage  110  is fed back  112  to an input  114 A of a phase-shift modulation circuit  114 . The modulation circuit provides four outputs  114 B connected at  116  to four inputs  118 A of a gate drive circuit  118 . The gate drive circuit  118  has four outputs  118 B connected at  120  to four inputs  104 C of the inverter  104 . 
     A number of power circuit configurations to implement the full-bridge inverter and resonant circuit of FIG. 1 are possible but the circuits as shown in FIGS. 2A and B are the circuits most commonly used in these implementations. 
     FIG. 2A shows the full-bridge inverter  104  and the resonant circuit  108  sections of a conventional inverter  200  which was described in ‘A 20 kHz Hybrid Resonant Power Source for the Space Station’,  IEEE Trans. on Aerospace and Electronics Systems,  vol. 25, No. Jul. 4, 1989, 491-496 by P. Jain &amp; M. Tanju. The full-bridge inverter  104  includes a first switch  202 , a second switch  204 , a third switch  206 , and a fourth switch  208 . Each switch  202 , 204 , 206 , 208  is preferably an N-channel field-effect transistor (FET). The resonant circuit  108  includes a series resonant circuit  210 , a parallel resonant circuit  212 , and a transformer  214 . 
     The full-bridge inverter  104  produces a quasi-square voltage at its output  106 , which is controlled using a phase-shift modulation circuit  114  (FIG. 1) commonly used in such applications. Both the series  210  and parallel  212  resonant circuits are tuned to an operating frequency of the inverter. Although the resonant circuit  108  produces a regulated sinusoidal voltage at its output  110 , this inverter  200  does not provide zero-voltage switching conditions for at least two of the four switches  202 , 204 , 206 , 208 , which results in higher switching losses at higher operating frequencies. Therefore, the operation of this circuit is limited to lower operating frequencies. 
     FIG. 2B shows the full-bridge inverter  104  and the resonant circuit  108  sections of a conventional inverter  250  which was described in ‘Constant frequency resonant DC/DC converter’, U.S. Pat. No. 5,157,593, Oct. 20, 1992 by P. Jain. The full-bridge inverter  104  is identical to the one shown in FIG.  2 A. The resonant circuit  108  includes a series resonant circuit  210 , a parallel resonant circuit  252 , and a transformer  214 . 
     The full-bridge circuit  104  produces a quasi-square voltage at its output  106 , which is controlled using a phase-shift modulation circuit  114  (FIG. 1) commonly used in such applications. In this configuration, the series resonant circuit  210  is tuned to an operating frequency of the inverter  250  while the parallel circuit  252  is tuned at a frequency, which is lower than the operating frequency. Although the resonant circuit  108  produces a regulated sinusoidal voltage at its output  110  and provides zerovoltage switching conditions for all the four switches  202 , 204 , 206 , 208 , the de-tuning of the parallel branch  252  requires the series resonant components  210  and the output transformer  214  to have higher maximum ratings and hence be more expensive. 
     Another fundamental problem that limits the operation of the inverter circuits of FIGS. 2A and B at higher operating frequencies is gate circuit losses of the FETs  202 , 204 , 206 , 208  used in the full-bridge circuit  104 . FIG. 3 shows a graph  300  of typical gating signals A 1   302 , A 2   304 , B 1   306 , and B 2   308  produced by the phase-shift circuit  114 . FIG. 4 shows a graph  400  of gate voltage (VgA 1 )  402 , gate current (igA 1 )  404 , instantaneous gate power (pgA 1 )  406 , and average gate power (PgA 1 )  408  for a gate  202 A of the first FET switch  202 . This graph  400  clearly shows that when a rectangular voltage pulse  402  is applied to the gate  202 A of the FET  202 , which has a capacitance, a pulsating current  404  is drawn from this voltage. This causes the power loss  406  in the gate circuit, which is approximately given by Cg*Vg 2 *f  408  (where Cg is gate capacitance; Vg is gate voltage; and, f is the operating frequency). At higher frequency, the gate losses are prohibitively high, which limits the operation of inverter circuits of FIGS. 2A and B at very high frequency. 
     It is clear from the above discussion that the conventional approaches to converting DC to high frequency AC have low conversion efficiency due to high switching losses. 
     There therefore exists a need for an inverter topology, which is capable of operating at substantially higher frequencies and has no, or very small, switching losses, including gate circuit losses. 
     SUMMARY OF THE INVENTION 
     It is therefore an object of the invention to provide a DC/AC inverter, which forms a high frequency sinusoidal AC source. 
     The invention therefore provides a high-frequency resonant sine wave DC to AC inverter suitable for use in a personal computer (PC) power supply, which includes a full-bridge inverter, a resonant circuit, a phase shift modulation circuit, and a resonant gate driver. The resonant gate driver provides sinusoidal gate drive signals to the full-bridge inverter enabling highly efficient operation on the inverter. 
     The invention further provides a method of driving an inverter to convert direct current (DC) to alternating current (AC), comprising a step of receiving square wave gating signals at a resonant gate driver and modifying the square wave gating signals to form sinusoidal gating signals for driving the inverter. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Further features and advantages of the present invention will become apparent from the following detailed description, taken in combination with the appended drawings, in which: 
     FIG. 1 is a block diagram of a conventional AC to high frequency AC voltage inverter; 
     FIGS. 2A and 2B are circuit diagrams of conventional full-bridge inverter and resonant circuits; 
     FIG. 3 is a graph of gating signals of a conventional DC to AC inverter; 
     FIG. 4 is a graph of gate voltage, current, instantaneous power, and average power of a gate of the inverter of FIG. 2A; 
     FIG. 5 is a block diagram of an AC to DC inverter in accordance with the present invention; 
     FIG. 6 is a graph of gating signals used in the AC to DC inverter of FIG. 5; 
     FIG. 7 is a graph of gate voltage, current, instantaneous power and average power of the inverter of FIG. 5; 
     FIG. 8 is a schematic diagram of a resonant gate drive circuit of FIG. 5; 
     FIG. 9 is a graph of voltage and current waveforms of the resonant gate drive circuit of FIG. 8; 
     FIGS. 10 to  12  are schematic diagrams of alternative embodiments the resonant gate drive circuit of FIG. 8; 
     FIG. 13A is a schematic diagram of the full-bridge inverter and resonant circuit of FIG. 5; 
     FIG. 13B is a graph of operating waveforms of the full-bridge inverter and resonant circuit of FIG. 13A; and 
     FIGS. 14 to  18  are schematic diagrams of alternative embodiments full-bridge inverter and resonant circuit of FIG.  5 . 
    
    
     It will be noted that throughout the appended drawings, like features are identified by like reference numerals. 
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     A block diagram  500  of a preferred embodiment of the present invention is shown in FIG.  5 . This block diagram  500  is identical the block diagram  100  shown in FIG. 1 except that the gate drive circuit  118  has been replaced by a resonant gate drive circuit  502 , and the resonant circuit  108  has been replaced by an improved resonant circuit  512  which provides loss-less switching of all the FETs  202 , 204 , 206 , 208  of the full-bridge inverter  104  without excessive rating of components in the resonant circuit  512 . These two aspects of the present invention are described below. 
     Resonant Gate Driver 
     The resonant gate driver  502  as shown in FIG. 5 consists of a conventional gate drive circuit  510  and a resonant circuit  504 . The gate drive circuit  510  has four inputs  510 A that are connected  116  to the gating signals  114 B (A 1 , A 2 , B 1 , B 2 ) from the phase-shift modulation circuit  114 . The gate drive circuit  510  generates rectangular voltage pulses  510 B (VgA 1   602 , VgA 2   604 , VgB 1   606 , VgB 2   608  in FIG. 6) that are connected  506  respectively to four inputs  504 A of the resonant circuit  504 . The resonant circuit  504  produces four sinusoidal voltage signals  504 B (VgA R1    610 , VgA R2    612 , VgB R1    614 , VgB R2    616  in FIG. 6) that are connected  120  respectively to the four inputs  104 C of the full-bridge inverter circuit  104 . 
     Now referring to the graphs  700  in FIG. 7, a brief description of the operation of the resonant gate drive  502  with respect to gate circuit losses is given here. For simplicity only gating signals for one gate are shown in FIG.  7 . Let us assume that the sinusoidal voltage  702  (Vg AR1 ) is produced by the resonant gate circuit  502 . This voltage  702 , when applied at the gate A 1 , produces a sinusoidal current igAR 1  at its output. Since the gate  202 A of the first FET switch S 1   202  is capacitive, current igA R1    704  is also sinusoidal but has a leading angle of 90° with respect to voltage VgA R1    702 . As a result an instantaneous power pgAR 1   706 , which is sinusoidal at twice the frequency of the gate voltage  702 , is drawn from the resonant gate drive circuit  502 . The instantaneous power pgA R1    706  has a zero average component (PgA R1 =0). This means the resonant gate circuit driver  502  results in a loss-less drive. (In actual practice, the average power is not ideally zero but has a small value due to the resistance associated with the components of the resonant gate driver circuit  510 . But this average power loss is significantly smaller than the Cg Vg 2  f losses of the conventional drive circuit  118 .) 
     A resonant gate drive circuit  800  in accordance with the present invention is shown in FIG.  8 . This circuit includes a gate driver  508 ; a series resonant circuit  802  comprising a series inductor  802 A having a value Lsg and a series capacitor  802 B having a value Csg; and a gate drive transformer  804  (Tg) having a primary winding  806  with Ni turns, a first secondary winding  808 , and a second secondary winding  810  each having N 2  turns. A parallel resonant circuit  812 , 814  comprising a parallel inductor  812 A, 814 A having a value Lg and a gate capacitor  812 B, 814 B having a value Cg. The series  802  and parallel  812 , 814  branches are tuned to a frequency of operation of the gate driver  508 . Now let us briefly explain the operation of the circuit  800  of FIG. 8 with the help of waveforms  900  as shown in FIG.  9 . 
     After receiving the signals  116  A 1  and A 2  from the phase-shift modulator  114  (PSM) on its input  508 A, the gate driver  508  generates a square-wave voltage  902  VgA at its output  508 B, the square-wave voltage  902  when applied at the series combination of the series resonant circuit  802  and primary winding  806  of the transformer  804  produces a sinusoidal voltage across the primary winding  806  of the transformer  804 . Since the parallel branch  812  is tuned to the operating frequency of the driver  508 , the application of the sinusoidal voltage across the parallel resonant circuit  812  produces two sinusoidal currents, iCg 1   906  through the capacitor  812 A and iLg 1   908  through the inductor  812 B. Both the currents  906 , 908  have the same magnitude but 180° phase difference. Similarly, the application of the sinusoidal voltage across the parallel resonant circuit  814  produces two sinusoidal currents, iCg 2   912  through the capacitor  814 B and iLg 2   914  through the inductor  814 A. Again, both the currents  912 , 914  have the same magnitude but 180° phase difference. The resultant currents ig 1   910  and ig 2   916  at the secondary windings  808 , 810  are, therefore, zero. This means the current drawn from the driver circuit  508  is also zero. 
     The above description reveals the following two characteristics of the resonant gate driver  502 : (1) the average power drawn from the resonant gate drive circuit  502  is zero, and (2) instantaneous current supplied by the gate driver  508  is zero. However, in actual practice, both the average power and current supplied by the driver  508  are not zero but have small values due to resistance associated with components of the resonant gate driver circuit  502 . An identical resonant gate drive circuit  800  as shown in FIG. 8 is used for driving gates  204 A, 206 A of the second switch  204  and third switch  206  of the fullbridge inverter  104  with the exception that the signals B 1  and B 2  are used as the input signals  116  instead of A 1  and A 2 . 
     FIG. 10 shows another embodiment of the resonant gate driver  1000  in accordance with the present invention in which a common parallel inductor  1002  having a value Lg is connected across the primary winding  806  of the transformer  804  and the inductors  812 A, 814 A across the secondary windings  808 , 810  are removed. 
     FIG. 11 shows another embodiment of the resonant gate driver  1100  in accordance with the present invention in which the parallel inductor  1002  of FIG. 10 is an integral part of the transformer  804 . 
     FIG. 12 shows another embodiment of the resonant gate driver  1200  in accordance with the present invention in which both the series  802 A and parallel inductor  1002  of FIG. 10 are integral parts of the transformer  804 . 
     Resonant Inverter 
     A DC/AC inverter in accordance with the invention is shown in FIG.  13 A and comprises a full-bridge inverter  104  comprising four switches  202 , 204 , 206 , 208 , a commutation inductor  1310 A having a value Lc, a blocking capacitor  1310 B having a value Cb, a high frequency transformer  214 , a series resonant circuit  210  comprising a series inductor  210 A having a value Ls and a series capacitor  210 B having a value Cs, and a parallel resonant circuit  212  comprising a parallel inductor  212 A having a value Lp and a parallel capacitor  212 B having a value Cp. 
     The full-bridge inverter  104  produces a quasi-square voltage at its output terminals. The commutation inductor Lc provides a zero voltage switching condition for the inverter switches in conjunction with the parallel capacitors  1302 , 1304 , 1306 , 1308  those are connected across the switches. The transformer T is used to match the output voltage level with the input voltage of the full-bridge. The components Ls and Cs of the series resonant circuit and the components Lp and Cp of the parallel resonant circuit are tuned at the operating frequency of the full-bridge inverter. Both the series and resonant circuits provide filtering, for the harmonics contained in the quasi-square wave of the full-bridge inverter, and produce a sinusoidal voltage output across the parallel resonant circuit. Capacitor Cb is used to prevent the saturation of the commutation inductor Lc. 
     A detailed description of the resonant inverter  1300  of FIG. 13A in conjunction with the operating waveforms  1350  as shown in FIG. 13B is now given. In operation when the resonant gate drive signals VgA R1 , VgA R2 , VgB R1 , VgB R2  are applied at the gates of switches  202 ,  204 ,  206 ,  208  respectively, a near quasi-square voltage waveform VAB  1362  is produced at the output  106  of the full-bridge inverter  104 . Since both the series  210  and parallel  212  resonant branches are tuned at the operating frequency of the inverter  1300 , a near sinusoidal current i s    1368  through the series branch  210 , a near sinusoidal voltage V p    1366  across the parallel branch  212 , and a trapezoidal current I LC    1370  through the commutation inductor  1310 A are established. For one cycle of operation of the inverter  1300 , the operation of the inverter  1300  is given below. 
     At time t=t 0 , only gate voltage VgB R2    1360  at the gate of the second switch  204  is above the gate threshold voltage VGth  1356 , which makes the second switch  204  continuously conduct. At the same time the net current iAB (is+ILC) is negative, which is forcing diode  1322  to conduct. 
     At t=t 1 , the gate voltage VgB R2    1360  falls below the threshold voltage VGth, the second switch  204  starts to turn-off and the negative current iAB starts to charge the second capacitor  1304  and discharge the third capacitor  1306 . By selecting the proper value of the second capacitor  1304 , the rate of rise of voltage across the second switch  204  can be controlled in such a way that the current flowing through the second switch  204  falls to zero before the voltage across the second switch  204  rises substantially. This results in near loss-less turn-off for the second switch  204 . 
     At t=t 2 , the second capacitor  1304  has charged to the level of input voltage Vi and the third capacitor  1306  has discharged to zero. The negative current iAB (is+ILC) now forces the third diode  1326  to conduct. 
     At t=t 3 , the gate voltage VgA R1    1352  rises above the gate threshold voltage VGth  1356 , the first switch  202  now starts to conduct. It should be noted that the first switch  202  turns-on under zero voltage as the first diode  1322  across it was conducting prior to the turn-on. 
     At t=t 5 , the gate voltage VgB R1    1358  rises above the gate threshold voltage VGth  1356 , the third switch  206  now starts to conduct. It should be noted that the third switch  206  turns-on under zero voltage as the third diode  1326  across it was conducting prior to the turn-on. 
     At t=t 6 , the gate voltage VgA R1    1352  falls below the threshold voltage VGth  1356 , the first switch  202  starts to turn-off and the positive current iAB starts to charge the first capacitor  1302  and discharge the fourth capacitor  1308 . By selecting the proper value of the first capacitor  1302 , the rate of rise of voltage across the first switch  202  can be controlled in such a way that the current flowing through the first switch  202  falls to zero before the voltage across the first switch  202  rises substantially. This results in near loss-less turn-off for the first switch  202 . 
     At t=t 7 , the first capacitor  1302  has charged to the level of input voltage Vi and the fourth capacitor  1308  has discharged to zero. The positive current iAB (is+ILC) now forces the fourth diode  1328  to conduct. 
     At time t=t 8 , only gate voltage VgB R1    1358  at the gate of the third switch  206  is above the gate threshold voltage VGth, which makes the third switch  206  continuously conduct. At the same time the net current iAB (is+ILC) is positive, which is forcing the fourth diode  1328  to conduct. 
     At t=t 9 , the gate voltage VgB R1    1358  falls below the threshold voltage VGth  1356 , the third switch  206  starts to turn-off and the positive current iAB starts to charge the third capacitor  1306  and discharge the second capacitor  1304 . By selecting the proper value of the third capacitor  1306 , the rate of rise of voltage across the third switch  206  can be controlled in such a way that the current flowing through the third switch  206  falls to zero before the voltage across the third switch  206  rises substantially. This results in near loss-less turn-off for the third switch  206 . 
     At t=t 10 , the third capacitor  1306  has charged to the level of input voltage Vi and the second capacitor  1304  has discharged to zero. The positive current iAB (is+ILC) now forces the second diode  1324  to conduct. 
     At t=t 11 , the gate voltage VgA R2    1354  rises above the gate threshold voltage VGth  1356 , the fourth switch  208  now starts to conduct. It should be noted that the fourth switch  208  turns-on under zero voltage as the fourth diode  1328  across it was conducting prior to the turn-on. 
     At t=t 12 , the gate voltage VgB R2  rises above the gate threshold voltage VGth, the second switch  204  now starts to conduct. It should be noted that the second switch  204  turns-on under zero voltage as the second diode  1304  across it was conducting prior to the turn-on. 
     At t=t 13 , the gate voltage VgA R2  falls below the threshold voltage VGth, the fourth switch  208  starts to turn-off and the negative current iAB starts to charge the fourth capacitor  1308  and discharge the first capacitor  1302 . By selecting the proper value of the fourth capacitor  1308 , the rate of rise of voltage across the fourth switch  208  can be controlled in such a way that the current flowing through the fourth switch  208  falls to zero before the voltage across the fourth switch  208  rises substantially. This results in near loss-less turn-off for the fourth switch  208 . 
     At t=t 14 , the fourth capacitor  1308  has charged to the level of input voltage Vi and the capacitor  1302  has discharged to zero. The negative current iAB (is+ILC) now forces the first diode  1322  to conduct. 
     At t=t 15 , a new cycle begins and the operation of the inverter  104  as described above repeats. 
     From the above description, it is clear that the switches of the inverter  104  are turned-on and turned-off with near zero switching losses. 
     Controlling the phase shift (φ) of the full-bridge inverter  104  controls the high frequency sinusoidal output voltage. 
     FIG. 14 shows another embodiment  1400  of the resonant inverter  500  OF FIG. 5 in which the parallel resonant circuit  1402  of the resonant circuit  512  is connected across the secondary winding of the transformer  214 . 
     FIG. 15 shows another embodiment of the resonant inverter  500  of FIG. 5 in which both the series  1502  and the parallel resonant  1402  circuits of the resonant circuit  512  are connected across the secondary winding of the transformer  214 . 
     FIG. 16 shows another embodiment of the resonant inverter  500  of FIG. 5 in which the parallel resonant inductor of the resonant circuit  512  is an integral part of the transformer  214 . 
     FIG. 17 shows another embodiment of the resonant inverter  500  of FIG. 5 in which the series resonant inductor of the resonant circuit  512  is an integral part of the transformer  214 . 
     FIG. 18 shows another embodiment of the resonant inverter  500  of FIG. 5 in which both the series and the parallel resonant inductors of the resonant circuit  512  are the integral parts of the transformer  214 . 
     Prototype Inverter System 
     A prototype of high frequency resonant inverter system of FIG. 15 was built to verify the performance. The inverter system is used to produce a 1 MHz, sinusoidal 28 Vrms, and 240 volt-ampere output power from an input voltage of 400 V DC. The following parameters are used for the power circuit: Cb=1 uF  1310 B, Lc=76 uH  1310 A, transformer  214  turns ratio (N 1 /N 2 =35/3), Ls 1 uH  1502 A, Cs=25 nF  1502 B, Lp=0.43 uH  1402 A, Cp=59 nF  1402 B, the switches  1302 , 1304 , 1306 , 1308  are IRF 840. The following parameters are used for the resonant gate driver  800  of FIG.  8 : Lsg=25 uH  802 A, Csg=1 nF  802 B, turns ratio for the gate transformer  804  (N 1 /N 2 =10/10), Lg=18 uH  812 A, 814 A, and Cg=1.3 nF  812 B, 814 B. The output voltage had lower than 1.5% total harmonic distortion, better than 1% voltage regulation and over 96% efficiency including the gate circuit. 
     The invention therefore provides an AC to DC inverter capable of operating at high frequencies and has very small switching losses. 
     The embodiment(s) of the invention described above is(are) intended to be exemplary only. The scope of the invention is therefore intended to be limited solely by the scope of the appended claims.