Abstract:
An integrated test core for mixed-signal circuits comprises a periodic waveform generator capable of generating arbitrary band-limited waveforms for excitation purposes and a waveform digitizer for extracting an arbitrary waveform from the test circuit&#39;s analog response signal. The digitized response may be tested and measured using DSP techniques. Preferably, the waveform generator and digitizer are synchronously controlled. The core is a nearly all digital implementation with the exception of a reconstruction filter (optional) for sending the test signal to the circuit under test (CUT) and the comparator for extracting the digitized waveform from the CUT&#39;s response. The periodic waveform generator may comprise a ΣΔmodulator and, optionally, a reconstruction filter between the modulator and CUT. The waveform digitizer may comprise a programmable reference voltage generator for providing a variable voltage reference signal, a comparator for generating a comparison signal from the CUT&#39;s analog response signal and the reference signal and means for controlling the reference voltage generator.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS  
       [0001]    This application claims the benefit of United States Provisional application Ser. No. 60/200,500 filed Apr. 28, 2000 and United States provisional application Ser. No. 60/205,468 filed May 19, 2000. 
     
    
     
       FIELD OF THE INVENTION  
         [0002]    The present invention relates to analog test and measurement apparatus, more specifically, it relates, to on-chip integrated excitation/extraction,systems of analog test a,nd measurement.  
         BACKGROUND OF THE INVENTION  
         [0003]    The combination of an increased demand for consumer electronics and the continued growth in semiconductor packing density is driving towards the integration of more and more system functionality onto a single integrated circuit (IC), The result, among other things, is an increased need for the integration of analog and mixed-mode components (eg. analog-digital, RF-analog-digital, and mechanical-analog-digital) on the same chip as the digital components, Designing such mixed-signal systems-on-chip (SOCs) is distinctively challenging since it entails managing system level abstractions while simultaneously dealing with physical effects at the transistor and parasitic levels, In the same manner, testing next-generation SOCs represents a real challenge, especially since cost and time-to-market are usually key requirements. Such mixed-signal ICs contain complex signal paths and functional specifications, and post-design ad hoc test program development will no longer be viable since it can significantly slow down device characterization and debugging, and it can tie up automatic test equipment (ATE) resources (in the production phase) and greatly increase the time-to-market.  
           [0004]    The difficulty is accentuated by another aspect of system-level integration, namely, the integration of third-party cores. In order to cope with design complexity, final system manufacturers are forced to rely on pre-designed blocks, or “cores,” and to integrate these cores as part of the bigger, more complex systems. These cores are obtained from virtual library (software) descriptions of the final IC block. In the digital domain, test access mechanisms (e.g. through scan) and design-for-test (DfT) techniques are already in place for the most part, and test information (digital bits) can be transported without loss throughout the SOC and across the chip boundary to the outside world. Thus, it seems possible to derive a systematic procedure by which the final system integrator can access the embedded digital virtual cores, The problem in the analog domain is the fact that it is a Jot harder to “scan” signals over long distances in a chip and across the chip boundary to the outside world. Rapid signal degradation due to digital noise coupling or analog buffer distortion characteristics is very likely to occur,  
           [0005]    In the present state of the art, analog and mixed-signal parts are tested externally for the most part. Even as core-based design grows in popularity,; the most critical high-frequency analog cores are still allocated dedicated and handcrafted I/O access so that they can be tested to specifications using external instruments, This being said, recent attempts at integrating some test functionality have been made, although they remain to be ad-hoc and customized in nature. For example, with reference to FIG. 1, an on-chip ramp generator for the code-density testing of A/D converters known in the art is illustrated. However, this technique is very specific to a certain class of AID converters. As well, a sine-wave generator:,using an oversampling delta-sigma oscillator has, recently been proposed (FIG. 2 a ). To. avoid the design complexity associated with such an oscillator, a circular memory based, approach that approximates the output of, a delta-sigma oscillator has been reported (FIG. 2 b ). This proposed design has the advantage of flexibility and the potential for a higher speed of operation. When both an AID and a D/A converter are present, prior art assumes the A/D can be tested somehow (e.g. using the circular memory approach), and then the A/D is used to test the, D/A, (FIG. 3). Beyond A/D and D/A converters, other techniques use an analog test bus to matrix signals around the chip to boundary elements containing switches, buffers, and/or comparators.  
           [0006]    As can, be seen, the above approaches are limited customized solutions or are generally cumbersome to implement in a virtual core-based, design environment. There is a need for a general integrated excitation/extraction system for analog test and measurement,  
         SUMMARY OF THE INVENTION  
         [0007]    According to one aspect of the invention, there is provided a reliable and compact integrated test system for analog and mixed-mode circuits. It consists of an almost all-digital implementation, which means that it benefits from advances in digital design and test (e.g. automatic synthesis and layout, and easier verification). It is capable of generating waveforms (to excite the analog circuit under test) and coherently and synchronously digitizing periodic analog waveforms (to extract information about the circuit under test) while utilizing an area that is only equivalent to about five thousand standard cell gates. It is stand-alone, in the sense that it does not require the availability of any specialized on-chip structures, and it does not require external analog signals. Through programming in software, this system can perform curve tracing, oscilloscope, and spectrum analyzer functions, all in one embedded core.  
           [0008]    According to a general aspect of the present invention, the on-chip integrated excitation/extraction system for analog test and measurement comprises a signal generator for generating a test signal to be sent to components of a circuit under test and a periodic waveform signal digitizer for analyzing a response signal from the components of the circuit under test. Preferably, the signal generator and the periodic waveform signal digitizer are synchronized using a single clock source, An analog reconstruction filter can be used to extract the analog signal for the CUT from the test signal bit sequence.  
           [0009]    According to another aspect of the invention, a possible digital access mechanism is described, in which the integrated test/measurement system is transformed into a serial scan register. As such, a multitude of such integrated test systems can be integrated onto the same SOC in order to excite and measure a multitude of embedded cores, Since the core responses are digitized locally, their responses can subsequently be moved around the IC (through the digital scan paths) without loss of information.  
           [0010]    According to a further aspect, a method for integrated excitation/extraction for test and measurement of a circuit under test is provided. In view of the above, it is the intent of this invention to provide a general test solution for analog cores that does not require the existence of D, or D/A converters. It is another intent to provide a solution for the in situ digitization of the CUT response signals and transporting test information in the digital domain, rather than in real-time in the analog domain. It is further our intent,to provide a multitude of these on-chip systems at various locations on an ICC to, measure analog effects at, multiple locations on a chip.  
           [0011]    Within the scope of this, invention, the definition of an analog core encompasses a digital core that has to be measured, to analog specifications. For example, digital signal rise/fall time or jitter are considered analog phenomena, as are digital signal coupling or digital switching noise. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0012]    Further features and advantages of the present invention will become apparent from the following detailed description taken in combination with the appended drawings, in which.  
         [0013]    FIG. 1 (Prior Art) is a schematic block diagram of an on-chip ramp generator for the code-density testing of A/D converters;  
         [0014]    FIGS. 2a and 2b (Prior Art) illustrate schematically a sine-wave generator using a delta-sigma oscillator and a memory based approach to approximate the output of a delta-sigma oscillator respectively;  
         [0015]    FIG. 3 (Prior Art) illustrates a test system for circuits having both an AID converter and a D/A converter;  
         [0016]    [0016]FIG. 4 shows functionally the components comprising the integrated excitation/extraction system for analog test and measurement of the present invention;  
         [0017]    [0017]FIG. 5 shows a more detailed view of the invention of FIG. 4;  
         [0018]    [0018]FIG. 6 shows a preferred implementation of signal generation components of the present invention of FIG. 5;  
         [0019]    [0019]FIG. 7 shows the power spectral density (PSD) of a typical finite-length approximation to a ΣΔ output;  
         [0020]    [0020]FIG. 8 illustrates sampling a ΣΔ modulation;  
         [0021]    [0021]FIGS. 9 a  and  9   b  illustrate the spectrum of a set of N consecutive ΣΔ outputs with a signal encoded at ˜F S /4, and a zoom showing the signal at 5 F S /4, respectively;  
         [0022]    [0022]FIGS. 10 a  and  10   b  show two signal encoding schemes;  
         [0023]    [0023]FIG. 11 illustrates the invention of FIG. 4 in a “self-test mode” where the output of the signal generator is directly connected to the signal digitizer;  
         [0024]    [0024]FIG. 12 shows an exemplary analog reconstruction filter;  
         [0025]    [0025]FIG. 9. shows test signal sub-sampling without modifying frequency content;  
         [0026]    [0026]FIG. 13 illustrates graphically a multiple-pass digitization comparison algorithm;  
         [0027]    [0027]FIG. 4 shows a flow-chart of the algorithm of FIG. 13;  
         [0028]    [0028]FIG. 15 graphically illustrates clocking to reduce incoming sample speed;  
         [0029]    [0029]FIGS. 16 a  and  16   b  show I/O mechanisms for the invention of FIG. 5;  
         [0030]    [0030]FIG. 17 illustrates a further embodiment of the invention of FIG. 5; and  
         [0031]    [0031]FIGS. 18 a  and  18   b  illustrate additional embodiments of the invention of FIG. 5. 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0032]    [0032]FIG. 4 shows a “functional” diagram of the proposed integrated excitation/extraction system for analog test and measurement, The system  100  comprises a means for generating arbitrary band-limited waveforms  105  and a means for synchronously digitizing  110  the periodic response of a circuits under test (CUT)  120 . A simple clocking system  115  synchronizes the two means  105 ,  110 . The output of the periodic waveform digitizer  110  is preferably stored in a digital memory  210  of FIG. 5 and may be connected to a digital signal processor (DSP)  125  that preferably resides on-chip for analyzing the digitized waveform. DSP  125  could be a general-purpose DSP that a system integrator is embedding in the final device, or it could be a specialized engine (like a Fast Fourier Transform computer or a digital filter), which could also be functional in the final product, A more detailed view of embedded test system  100  is shown in FIG. 5.  
         [0033]    With reference to FIG. 5, system  100  preferably comprises first and second one-bit memory periodic bit-stream generators  217 ,  215  preferably implemented as sequential shift registers as will be described further below, analog structures  204 ,  206 , and  209  and a multi-bit memory  210  for output data storage (parts of which memory could be implemented externally to the device). A key feature of this architecture is its simplicity and modularity. First and second one-bit memory  200 ,  202  implement two periodic bit stream generators  217  and  215 . First one-bit memory  200  provides arbitrary AC signal generation for exciting CUT  120  while second one-bit memory  202  provides DC signal generation. As was described earlier, a short bit sequence can be chosen to encode a high-quality periodic analog signal (e.g. a sine wave). When this sequence is periodically repeated and passed through a relatively imprecise analog reconstruction filter  230 , a high quality analog signal can be generated, Thus, first one-bit memory  202  is used to synthesize periodic analog signals in accordance with this technique.  
         [0034]    Second one-bit memory  202  is used with averaging circuit  204  to encode high-precision DC levels that sweep the whole range of the device supply voltages. These DC voltages are combined by an analog comparator  206  to perform arbitrary amplitude resolution digitization. An algorithmic progressive AND conversion operation that relies on the periodicity of the CUT response signal is utilized and described further below with reference to FIGS.  13 - 15 .  
         [0035]    [0035]FIG. 5 also shows a straightforward implementation of the clocking system  115 . For reasons that will be apparent later, first and second one-bit memory  200 ,  202  are expected to run at or close to the maximum speed of the technology, but the same cannot be said about comparator  206  (and hence the output memory  210 ). In a preferred implementation, clocking system  115  implements a frequency divider  116  to slow down the incoming samples to comparator  206 , while not compromising the effective sample rate of digitizer  110 .  
       Periodic Bit Stream Generators  
       [0036]    In monolithic form, the periodic bit stream generators  217 ,  215  (comprising respectively first and second one-bit memories  202 ,  204 ) can simply be implemented in the manner shown in FIG. 6. In this figure, daisy-chained D Flip Flops  231 ,  233  are used and a control signal  235  determines whether to serially load the chains from data input  237  or to periodically circulate their contents. Both memories  200 ,  202  may be loaded through the same input port  237  through a simple multiplexing mechanism, This implementation is attractive because it is extremely easy to synthesize and can benefit from very high speeds of operation. The D Flip Flops  231 ,  233  should be optimized for speed and low power dissipation. It should be noted, however, that many forms of embedded digital “memory” could be used to perform the periodic circulation of digital bits.  
       Encoding an AC Signal  
       [0037]    A short bit sequence may be chosen to approximate the output of a one-bit sigma-delta (ΣΔ) modulator when driven by a periodic signal. The concept behind this approach is described in the art and an application entitled “Programmable DC Voltage Generator” assigned to the owner herein and incorporated herein by reference and the concept consists of simulating a high-order noise shaping modulator and capturing a finite duration (finite number of samples) of its output. Periodically repeating this finite segment approximates the usually chaotic output of the ΣΔ modulator. However, in order for the periodic approximation to achieve a high fidelity, the frequency of the input signal to the ΣΔ modulator has to be harmonically related to the fundamental frequency of the bit sequence: if N is the length of the approximate ΣΔ sequence and F S  is the sampling rate, then, the input to the modulator has to be a multiple of F S /N. In fact, the forced periodicity of the approximate ΣΔ output also means that the encoded signal contains only multiples of this same fundamental frequency:  
                 f   out     =       M   N          F   s         ,     M   =   0     ,   1   ,   2   ,   Λ   ,     N   2             (   1   )                               
 
         [0038]    This is a very important feature of the bit stream generator that makes it favorable over other signal generation techniques, described earlier. Specifically, it guarantees sample coherence with our on-chip circuit-response digitizer. Coherent sampling enables the use of a small number of samples in a DSP-based measurement environment. FIG. 7 shows the power spectral density (PSD) of a periodic At stream. As can be seen, as long as the encoded signal lies within the modulator bandwidth, it will have very low distortion harmonics. The encoded signal in this case is a multi-tone signal, which is typically used for frequency response measurements.  
         [0039]    It is important to note that randomly choosing a set of N consecutive outputs of the ΣΔ modulator does not generally produce, a signal having the high quality that is depicted in FIG. 7. The reason is that, as discussed above, the output of a ΣΔ modulator is generally a periodic even if its input is periodic. Consequently, only an infinitely long sequence of bits is expected to approach the true ΣΔ modulator output&#39;s, dynamic range and noise-shaping properties. Instead, some form of optimization is needed in the choice of the NΣΔ outputs. Selecting an optimum bit stream has been, described earlier in the literature and is thus beyond the, scope of this patent. In a nutshell, and with reference to FIG. 8, the ΣΔ modulator is simulated for a very long time duration and sets of N consecutive bits are captured and analyzed, The algorithm continues to select different sequences (of length N) until the desired spectral properties are achieved.  
         [0040]    In the prior art, the periodic bit stream has been used to generate signals that are located within the Nyquist rate of F S /2. In this application, since the on-chip digitizer has the potential of capturing much higher test bandwidths, there is an option of using the signal images that arise in the frequency domain as a result of periodically repeating the bit sequence. Specifically, the spectrum of the periodic signals generated using this method will have copies at multiples of F S , although the energy levels at increasing frequencies keep decreasing and eventually, they cannot be resolved from the thermal noise. FIG. 9 shows an example where a signal that is encoded at approximately F S /4 using the above technique also appears at about 5F S /4 and 7F S /4. Provided these tones can be filtered and detected by the circuit under test, the on-chip digitizer can also be used to capture responses at these frequencies.  
       Encoding a DC Signal  
       [0041]    As will be described shortly, arbitrary precision digitization is achieved using only a single analog comparator  206  with varying values of the reference input  208  to the comparator  206 . These DC reference levels need to be generated accurately over the whole range of input amplitudes, since their linearity directly influences the overall linearity of the complete capture system. To achieve these requirements, digital pulse modulation techniques are used, in which the desired DC level is encoded in the average of a periodic digital sequence. This has the advantage of an almost guaranteed linearity (assuming the digital electronics function properly).  
         [0042]    The encoding scheme for the reference level is pulse-density modulation, which also relies on over-sampling, noise-shaping principles. The main reason for this is to combat the shortfalls of traditional parallel-serial conversion techniques that possess high linearity for low-cost DC (low frequency) D/A converters like pulse-width modulation (PWM). Pulse-width modulation has poor spectral properties, which renders it impractical in an application like this one. For example, the rectangular waveform of FIG. 10( a ), which encodes a DC level of 0.75 V 00 , and whose Fourier Series can be expressed as  
               x        (   t   )       =         768   1024          V   DD       +       V   DD            ∑     k   ≠   0              1     k                 π            sin        (       768   1024        k                 π     )                 j                 k          2                 π                   F   s       1024            f                ′                         (   2   )                               
 
         [0043]    has most of its harmonic energy concentrated at a frequency of F S /1024, a very low frequency. A filter that sufficiently attenuates this harmonic energy will need to have a rather big time-constant, and the latter has to increase as the number of encoded levels, N B , increases. With ΣΔ modulated streams, which are conceptually illustrated in FIG. 1  0 (b), we can encode the same DC levels while benefiting greatly from the frequency characteristics of pulse-density modulation (PDM). For example, the Fourier Series representation of the signal of FIG. 10( b ) is given by  
               x        (   t   )       =         3   4          V   DD       +       V   DD            ∑     k   ≠   0              1     k                 π            sin        (       3   4        k                 π     )                   j                 k          2                 π                   F   s       4        f                                    (   3   )                               
 
         [0044]    where the dominant harmonic now sits at a frequency of F S /4, a much higher frequency A low-pass filter that attenuates this harmonic to the required ripple magnitude will have a much smaller time-constant (hence smaller implementation area) than one that would be used in the case of PWM sequences. In general, ΣΔ representations of arbitrary DC signals will still contain harmonics at F S /N B . However, noise shaping properties of software ΣΔ modulators ensure that these will be low enough in power so as not to dominate the design of the averaging circuit  204  (e.g. a low-pass filter).  
       Analog Filtering  
       [0045]    Two analog averaging or filtering operations are performed, one for the AC generator  200 , and one for the DC generator  202 . Concerning the first one bit memory  200 , a filter is generally required to reconstruct the analog signal, and the type of filter will ultimately depend on the application of interest. In some applications, the filter is part of the CUT  120  as illustrated in FIG. 5. For example, the CUT might itself be a filter (low pass or band pass), or it might perform a more complex function that is preceded by a filtering operation. If no filter is present, then a relatively imprecise filter has to be included. The design of the filter is facilitated by the noise-shaping properties of ΣΔ modulation, which means that the filter needs only be tuned to the pass-band of the ΣΔ stream, regardless of the encoded analog signal. More importantly, since AC bit stream generator  200  is fully programmable, the architecture of the present invention is not restricted to a particular modulator order or topology. Thus, the spectral properties of the programmable ΣΔ streams can be shaped in such a manner that compensates for the roll-off behavior of the implemented filter within the desired test-system bandwidth. Specifically, the implemented filter response can be measured using integrated capture system  100  and then compensated for in the bit stream generation phase. Consider a situation in which the test system is connected in the manner shown in FIG. 11, where the output of reconstruction filter  230  is directly connected to waveform digitizer  110 . In this configuration, a multi-tone signal whose frequency content is chosen to match the frequency response of filter  230  is encoded by periodic bit stream generator  200 . The stream is passed through filter  230  and the resulting waveform is captured using the on-chip waveform digitizer  110 . A FFT-based analysis can then be used to characterize filter  230 . Other types of tests can also be performed on filter  230  before overall test system  100  can be used to verify the other integrated analog circuits. For example, a model of the non-linearity introduced by arbitrary waveform generator  105  can, be created, and its effects can be cancelled in software when, say, the non-linearity of the circuit under test  120 , is being verified.  
         [0046]    Preferably, analog reconstruction filter  230  is an active RC filter. This type of filter has the advantage of providing very good linearity and dynamic range and meets the attenuation specifications for ΣΔ-encoded bit streams. Moreover, it is relatively easy to integrate in monolithic form. FIG. 12 shows a single-ended example of a 4 th -order low-pass filter, although a 4 th -order filter is not required. As was mentioned earlier, the final filter requirements will ultimately be dictated by the application. In other words, there is a compromise between the noise-shaping properties of the generated ΣΔ bit streams, and the roll-off requirements of analog reconstruction filter  205 . Moreover, the fact that the roll-off behavior of filter  205  may be measured means less stringent requirements on the filter transition band can be tolerated, without significantly compromising the usable bandwidth for test tone generation.  
         [0047]    As for the DC generator  202 , the average of the periodic bit stream is extracted using only a passive RC filter. The reason is that such a filter yields the DC component of the input signal with maximum accuracy, which is key to achieving robustness to process variation. An examination of the transfer function for the first order case reveals that the gain at DC is unity regardless of the value of RC:  
                    H        (   f   )            =     1       1   +       (     2      π                 fRC     )     2                   (   4   )                               
 
         [0048]    In practice, higher order filters are desirable since they have the potential of achieving a smaller time-constant for the same ripple requirements (i.e. bandwidth). A passive RC filter is also compact and benefits directly from scaling as the modulation frequency increases with process improvements. In fact, the filter response of (4) and, similarly, the filter response for the AC generator are the main reasons why first and second one-bit memories  200 ,  202  of FIG. 5 are preferably circulated at the maximum speed of the technology. A faster clock rate means higher frequency signals and hence smaller filter implementation areas and faster settling times. Running at the maximum speed of the technology is possible since there are no logic elements between the Flip Flops  231 ,  233  in FIG. 6. For example, it is not unreasonable to achieve a rate of 1 GHz in a 0.25 μm standard digital CMOS process,  
       Integrated Circuit Response Capture  
       [0049]    The combination of comparator  206  and the DC bit stream via, input  208  enables us to achieve arbitrary analog waveform digitization. The periodicity of the analog signal under test (which, is enforced in the present architecture) enables: one to make multiple comparison passes over progressive periods of this signal.  
         [0050]    The way the signal is digitized is as follows. Assume that the number of samples in the unit test period is N. and the number of quantization levels in the overall A/D conversion operation is 2 M  (according to FIG. 5, this is also equal to A. First, programmable reference voltage generator  215  is commanded to output the lowest possible quantization level. All samples of the unit test period are collected and compared to this quantization level, Once all comparisons are made (stored in memory), reference voltage generator  215  is commanded to increment its output to the next quantization level, and the process repeats. Specifically, since the analog waveform to be digitized is periodic, the quantization level can be incremented and sampling delayed for as long as desired before collecting new samples for comparison (as long as the start of the new comparison pass is at the same location on the unit test period). FIG. 13 illustrates this multiple-pass comparison algorithm graphically, and FIG. 14 shows a flowchart of this procedure As can be seen in FIG. 13, since the waveform is to be quantized using 16 levels, the conversion process covers  16  cycles of the waveform. In each cycle, all N waveform samples are compared to the respective quantization level. Realistically, however, the programmable reference voltage level cannot be incremented instantaneously, So, after each comparison pass, at least one waveform cycle has to pass (in order to allow the output level to settle to its final value) before the next comparison pass can be performed.  
         [0051]    As can be seen, our capture algorithm relies primarily on the, proper synchronization with the excitation system, Specifically, in each comparison pass the comparator is expected to see the exact same samples of the test signal as it did in the previous passes. This is easily achieved using the bit stream generation approach, since all the generated tones are well defined with respect to the sampling clock,system  115 .  
         [0052]    Similarly, the clock speed that the periodic bit stream generators  200 ,  202  run at is generally faster than the comparison speed of typical integrated voltage comparators (e.g.  206 ). Under these circumstances, a sample-and-hold circuit  205  (which can track and sample very fast signals) can be inserted at the corresponding input of comparator  206  to receive the signal from circuit under test  120 . Sample-and-hold circuit  205  introduces a means for sub-sampling at another level for each quantization level as demonstrated graphically in FIG. 1Sa and FIG. 1Sb. In this figure, comparator  206  is assumed to take as many as three clock cycles to arrive at a decision about the relative magnitudes of its two input signals. To accommodate comparator  206  while still achieving a high effective sampling rate, sample-and-hold circuit  205  can be clocked in the manner shown in FIG. 15 a.  In this example, if the periodic input signal is allowed to cycle three times, all samples of this period will be collected. So, in effect, each comparison pass in FIG. 13 will now take three unit test periods instead of just one. In general, with additional reference to FIG. 5, the ratio of the bit stream clock  114  and the comparator, clock  117  has to be a prime relative to the total number of samples,,in a test period, N, which is the same as the requirement for coherent sampling (described further below). For example, in FIG. 15 (since N is a power of 2), the comparator can take 5 or (7) clock cycles to arrive at a comparison decision, but 5 (or 7) periods of the input signal have to complete over each comparison pass.  
         [0053]    It should be noted that the means for sub-sampling in this application also encompasses a passive sampling network (consisting only of switches and capacitors) at the front-end of a comparator. Full-featured sample-and-hold circuits (e.g. with input and output buffers or feedback loops) are not needed in such a case since output of the sample-and-hold is fed directly to a comparator at a well-defined time instant. In some occasions, fast sample-and-hold circuits (even passive ones) can exhibit a small amount of non-linearity that might affect the linearity of the complete measurement system. To avoid this, another means for sub-sampling (e.g. sample-and-hold circuit  207 ) is placed at the negative input  208  of the comparator  206 . This way, the two paths at the input of comparator  206  are identical (assuming matched devices and boundary conditions). Provided that sample-and-hold circuits  205  and  207  are monotonic, the difference voltage between the two inputs of comparator  206  is not expected to change polarity due to non-linearity of the sample-and-hold circuits  205 ,  207 .  
         [0054]    Processing of the comparator output is done using a multi-bit memory  210  (FIG. 5) that is preferably the same length as the length of the unit test period and that is initialized to zero at the beginning of a measurement run, For each comparison pass, the bit stream generators  217 ,  215  continuously circulate their contents to output the analog stimulus and the reference level, respectively, For each circuit response sample, the corresponding memory location is incremented or left unchanged, depending on whether the comparator output (for the current DC level) was 1 or 0. Thus, the adder  209  in FIG. 5 is just an increment-by-one structure that increments the memory location depending on the comparator output. At the end of the digitization process, each memory location contains an integer count representing the quantization level for the corresponding sample (ire. a thermometer code). Alternatively, the output memory  210  can have a width of 1 bit, and the increment operation can be performed externally to the chip.  
       Clocking System  
       [0055]    As was mentioned earlier, in its simplest form, synchronous control clocking system  115  provides a simple integer frequency divider  116  (which can be implemented as a counter) in order to accommodate the lower speed of the on-chip voltage comparator. Integer divider  116  is chosen to set the comparator sampling rate in such a way that enables the digitization of the input test signal where all N samples appear to have been taken at evenly distributed times over one cycle of the test signal, To that end, the conditions for coherent sampling, have to be satisfied. Specifically, N, the number of samples in the periodic test signal (determined by the bit stream lengths on the stimulus side) should be relatively prime with respect to DIV, the clock division ratio. FIG. 15 a  already illustrated the example of N=8 and DIV=3. Again, recall that the test signal is now circulated DIV times in order for all N samples to be stored. It should also be noted that this configuration allows for sampling signals at multiples of F S /N (refer to equation 1) even if these extend well beyond F S  (as is the case when images of the bit stream spectra are used). In other words, the effective bandwidth of the system is determined by the tracking bandwidth of the comparator input sampling network (e.g. sample-and-hold  205 ) and not by the division ratio, DIV.  
         [0056]    Some high-speed circuit phenomena occur at a periodicity that is associated with the overall system clock (e.g. digital switching noise in the power, supply rails or in the substrate) and/or are broadband in nature. In such applications, an, integer DIV does not suffice since phenomena at multiples of F S  create an ambiguity when aliased back. to the Nyquist interval (baseband); and since aliasing masks signal information in the case of broadband signals. A fractional frequency divider can be used for the sampling clock in order to provide for a much finer sampling resolution, but it is relatively hard to implement in monolithic form. Alternatively, digitizer sampling control  117  can be provided externally, so that test system  100  expects two clock frequency inputs (not shown). In this case, the periodic bit stream generators are circulated to perform a synchronization function to periodically trigger the on-chip event being measured, and the digitizer clock is used to sample the CUT response and progressively digitize it (over multiple runs) at a multitude of time instants within the synchronization clock period.  
       Access Mechanism  
       [0057]    [0057]FIG. 6 illustrates the aggregation of the first and second one-bit memories  200 ,  202  into a single scan chain. By a similar mechanism, the output memory can also be configured in such a manner. This way, the whole test system  100  may be transformed into a single scan chain. As system  100  is incorporated around a certain analog or mixed-signal block  120 , the usual core I/O are augmented with scan I/O. FIG. 16 illustrates this principle. This architecture is similar to digital core wrappers for standardizing access to embedded virtual cores, Multiple test systems may be configured similarly so that they resemble multiple scannable cores on the same chip.  
         [0058]    In the preferred embodiment of FIG. 4 and FIG. 5, CUT  120  is a conventional analog or mixed-signal component (e.g. analog filter, amplifier, or data converter). However, the present invention is also envisioned as an on-chip instrument that provides the capability to sample high-speed phenomena at multiple locations on a chip. For example, it was already mentioned that digital signal rise/fall times or serial communication buffers are considered within the realm of devices that can be measured using this test system. Such phenomena are considered analog ones. Similarly, digital signal coupling effects are increasing in importance as on-chip interconnection lengths increase and wire pitch decreases. Such effects can threaten the delivery of “digital” bits over long distances without corruption (hence device failure). In such a case, the proposed test system can also be used as illustrated in FIG. 17 (especially in the characterization and debugging phase). In this figure, the stimulus side is simply used as a synchronization signal to trigger a coupling event onto a victim wire  170  (the CUT in this case). The capture side is located at the desired “probe” location  175  on the wire, and the resulting voltage spike due to coupling from the aggressor wire  180  is measured. Capacitive and inductive parasitics can be deduced from the line response as is done, say, at the printed circuit board level conventionally. Note that the stimulus and capture sides in FIG. 17 could also be the respective components of two instances of test system  100  residing at the respective locations on-chip. Communication between the two systems is established in the digital domain, and the two systems are loaded and configured simultaneously according to the access mechanism described above. Reverse cross-talk (signal coupling as it affects the source of victim line  170 ) can also be measured by moving the probe point  175  to the beginning of victim line  170 , Similarly, the quality of digital signal transmission through a single on-chip,wire can also be evaluated according to a connection similar to FIG. 17 in which the input of comparator  206  is connected to the switching line itself, rather than a neighboring one (not shown).  
         [0059]    Similarly, CUT  120  can simply be a digital supply rail (or a ground substrate point) as illustrated in FIGS. 18 a  and  18   b.  Again, a trigger signal is periodically generated and the integrated digitizer is used to capture the voltage “bounce” (noise) that results due to the high switching currents of digital circuits. It is assumed in this configuration that the comparator  206 , which is the most critical component, is connected to a relatively clean supply, thus being less susceptible to the digitally induced noise.  
         [0060]    The present invention is useful in many environments and for many purposes, The invention may be useful as a field tester at the system level, thus providing potential for service cost reductions as field serviceability is becoming a significant portion in many industries, Moreover, the invention may be useful as a “tester” for a printed-circuit-board system. If a component on the board is suspected to have failed (after, for example, years of operation), the tester IC, could simply be programmed (locally for remotely) to excite the failing component and analyze its output. If the component does indeed fail the test, the component may then be replaced as applicable.  
         [0061]    The embodiment(s) of the invention described above is (are) intended to be exemplary only. The scope of the invention is therefore intended to be limited solely by the scope of the appended claims.