Abstract:
Systems and methods for avoiding non-linear behavior of conventional early-minus-late correlator-based code loop discriminators. The present invention provides a model of correlator behavior that takes into account whether both correlators are on the same side of the correlation peak or they straddle the peak. The result is a piecewise solution that can be stitched together quite readily to produce an extended range of linear response, thereby improving the pull-in capability of GPS code loops when closely spaced correlators are being used.

Description:
BACKGROUND OF THE INVENTION 
     In GPS receivers, pull-in performance refers to the ability of a tracking loop to converge to the desired state. A code loop can achieve the best pull-in performance when a code-loop discriminator output is linearly related to the temporal shift between the arriving code and the replica code produced by a local code generator. In conventional GPS receivers with early and late correlators separated by the width of a code chip, linear behavior in the discriminator can be achieved over an operating range of +/−½ the width of a code chip. However, it is widely recognized that better overall performance of the receiver can be achieved when the spacing between the early and late correlators are reduced. Such arrangements are particularly effective in reducing the impact of multi path interference, but it comes at a cost. As the separation between the early and late correlators is reduced below one chip width, the interval over which the discriminator behaves linearly also decreases. While this may not be a problem when the tracking loop is operating near the correct solution, performance during acquisition and the early stages of tracking can suffer, with degraded pull-in performance. 
     Conventional GPS receiver design involves a code-tracking loop whose job is to maintain temporal alignment between the arriving coarse/acquisition (C/A) code and the replica code produced by the local code generator. This is accomplished with the help of output from two correlators. One correlator employs a locally generated code sequence that is advanced a fraction of a chip interval relative to the prompt position, and is known as the early correlator. The second correlator operates with the locally generated code sequence delayed a fraction of a chip interval relative to the prompt position, and is known as the late correlator. Ideally, the correlation function exhibits triangular behavior (See  FIG. 1 ) with a peak at the prompt position and it decreases linearly as a function of the misalignment, out to a misalignment of one chip interval on either side of the peak. This behavior is illustrated in the following. 
     
       
         
           
             
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     Let CE represent the output of the early correlator and CL the output of the late correlator. The conventional method then considers these two correlator values to be located on opposite sides of the correlation peak. In that case:
 
 CE ( x,Δ,ampl )= ampl*cor ( x−Δ/ 2);
 
 CL ( x,Δ,ampl )= ampl*cor ( x+Δ/ 2);
 
     where Δ represents the displacement between the early and late correlators (i.e. temporal delay between two selected correlator values). The early and late correlators are assumed to be equally spaced by the amount Δ/2 on either side of the prompt correlator. 
     It is assumed that CE is to the left of the peak and CL is to the right. Under these conditions:
 
 CE=ampl *(1−Δ/2+ x )
 
 CL=ampl *(1−Δ/2− x )
 
     These equations can be solved for x to obtain 
     
       
         
           
             
               
                 
                   
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     with ampl having been eliminated. The subscript 2 is used to distinguish this particular solution from alternative solutions that follow. 
     Technically, this solution is only valid when |x 2 |≦Δ/2, but this equation is commonly used as a discriminator in the code loop without regard for this limitation. The case where Δ is set equal to the width of a code chip interval then becomes:
 
Δ=1;
 
 x   2 ( CE ( x,Δ,ampl ),  CL ( x,Δ,ampl ), Δ)= x  for | x|≦Δ/ 2=½
 
     As desired, x 2  behaves as a linear function of x over the range +/−(Δ/2). This means that the discriminator is providing an accurate estimate of the amount of misalignment between the arriving C/A code and the local code generator over this range. Outside this range, it no longer provides this linear response. Instead, it saturates at a value of +/−Δ/2. 
     When Δ is reduced to only 0.2 of the code chip interval, then this solution is only valid for |x|≦Δ/2=0.1 chip intervals. In the central region between −0.1 and 0.1, the discriminator behaves linearly, just as it did when Δ was set to 1. However, outside this range the curve increases more slowly than it would have if the linear behavior had continued. (See  FIG. 2 ). That means for displacements between Δ/2 and 1−(Δ/2), the discriminator value will be smaller than the actual misalignment amount. As a result, the code loop will make smaller corrections than it should, and hence the loop will not reach the desired tracking position as quickly. As a result, the pull-in behavior of the loop is degraded. The curve eventually reaches a value of 0.9 for a position on the horizontal axis equal to 0.9. This is larger than the saturation value obtained above when Δ was set to 1. So, for large misalignments greater than about 0.8 code chip intervals, the loop response will be better in this latter case than in the former. 
     Therefore, there exists a need for the linear behavior of the discriminator to extend over a greater range when the early to late correlator spacing Δ is on the order of 0.2 or 0.1 code chip intervals thereby improving pull-in performance. 
     SUMMARY OF THE INVENTION 
     The present invention provides systems and methods for avoiding non-linear behavior of conventional early-minus-late correlator-based code loop discriminators. The present invention provides a model of correlator behavior that takes into account whether both correlators are on the same side of the correlation peak or whether they straddle the peak. The result is a piecewise solution that can be stitched together quite readily to produce an extended range of linear response, thereby improving the pull-in capability of GPS code loops when closely spaced correlators are being used. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Preferred and alternative embodiments of the present invention are described in detail below with reference to the following drawings: 
         FIGS. 1 and 2  are plots as defined by the prior art; 
         FIG. 3  is a schematic drawing of a GPS receiver formed in accordance with an embodiment of the present invention; 
         FIG. 4  is a schematic drawing of signal processor used in the GPS receiver shown in  FIG. 3 ; and 
         FIGS. 5-8  are graphs of equation results produced by the signal processor. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
       FIG. 3  illustrates a block diagram of a GPS receiver  100  in accordance with an embodiment of the invention. It should be readily apparent to those of ordinary skill in the art that the block diagram depicted in  FIG. 3  represents a generalized schematic illustration and that other components may be added or existing components may be removed or modified. Moreover, the GPS receiver  100  may be implemented using various components. These components may be software components, hardware components, firmware, or any combination thereof. 
     As shown in  FIG. 3 , the GPS receiver  100  includes a pre-amp module  102 , a down conversion module  104 , a reference oscillator module  106 , a frequency synthesizer module  108 , an intermediate frequency (IF) demodulator  110 , a signal processing module  112  and a navigation module  114 . 
     A GPS signal is received at an antenna  116  and filtered by the pre-amp module  102 . The filtered RF signal is then down-converted by the down conversion module  104 . The down conversion module  104  is driven by a first local oscillator signal supplied by the frequency synthesizer module  108 , which itself is driven by the reference oscillator  106 . In addition, the frequency synthesizer module  108  provides a second local oscillator signal and a clock signal to the IF demodulator  110 . The IF demodulator  110  then generates digital in-phase and quadrature (I &amp; Q) channels by multiplication by sine and cosine versions of the second local oscillator signal. 
     The signal processing module  112  is configured to include a carrier loop and a code tracking loop (not shown). The signal processing module  112  also produces pseudorange, Doppler, and carrier phase outputs, which are then processed by the navigation processing module  114  to determine the receiver&#39;s position, velocity, and time. 
       FIG. 4  illustrates a more detailed block diagram of the signal processing module  112  shown in  FIG. 3  in accordance with another embodiment of the invention. As noted above, in order to track an incoming GPS signal, the signal processing module  112  employs two tracking loops—a delay lock loop (DLL)  200  and a phase lock loop (PLL). The DLL  200  is used to track the C/A code. The PLL (not shown) is used to track the carrier frequency of the incoming GPS signal. The DLL  200  includes a local code generator  202  and a code numerically controlled oscillator (NCO)  204 . 
     The code generator  202  generates three different phases of local replica codes—Early, Prompt, and Late, which are separated from adjacent neighbors by at most ½ of a chip. The early code is advanced with respect to the prompt code and the late code is delayed with respect to the prompt code. At each instant in time, 6 products are produced: early code x I, early code x Q, prompt code x I, prompt code x Q, late code x I and late code x Q. Each of these six values is added respectively to one of six integrators until it is time to dump the result. The result is 6 correlator values. 
     The integrate and dump module  208  sums the squares of the I and Q correlator values for each of the early, prompt, and late correlators. Then, the sums are square rooted to produce correlator values, CE, CP, CL. The correlator values are then provided to the code discriminator  210 . 
     The code discriminator  210  calculates an output by comparing the correlator values. The output is then fed back to the code NCO  204  to generate a feedback signal to the code generator  202 . The code generator  202  may then adjust its rate or phase based on the feedback signal. If observations provide values that are directly related to the amount of correction that is needed (as is the case with our discriminator and its improved linearity of response), then convergence (pull in) is faster and better behaved than is the case when the relationship between the observation and the correction departs from linearity. 
     Each of the three pairwise combinations of CE, CP and CL could be used to produce three discriminator results that could be combined in an appropriate manner to arrive at a final result. But only one pairwise combination is required to obtain a valid solution. Common practice has been to rely on the (CE,CL) pair. 
     Assume that both the early and late correlators are placed in positions to the left of the correlation peak. Under these conditions, the following expressions apply:
 
 CE=ampl *(1−Δ/2+ x )   (2)
 
 CL=ampl *(1+Δ/2+ x )   (3)
 
     CE is the same as before, but notice the change in the signs in the CL term. Equations 2 and 3 are solved for x in terms of CE, CL and Δ (ampl again drops out), to obtain: 
                       x   3     ⁡     (       C   ⁢           ⁢   E     ,     C   ⁢           ⁢   L     ,   Δ     )       =       -   1     -       Δ   /   2     ⁢         C   ⁢           ⁢   E     +     C   ⁢           ⁢   L           C   ⁢           ⁢   E     -     C   ⁢           ⁢   L                     (   4   )               
where the subscript 3 is used to distinguish this solution from the previous one denoted by a subscript of 2.
 
     This result is valid when both correlators are located to the left of the correlation peak, or equivalently whenever −1+Δ/2≦x≦1−Δ/2. The difference between the early and late correlator values appears in the denominator and the sum in the numerator. Only Δ/2 is multiplied by the ratio of correlator terms; the leading −1 term stands by itself. 
     This same process is repeated for the case where the early and late correlators are both to the right of the correlation peak, in which case the expressions become:
 
 CE=ampl *(1+Δ/2− x )   (5)
 
 CL=ampl *(1−Δ/2− x )   (6)
 
     In this case, the solution takes the form: 
     
       
         
           
             
               
                 
                   
                     
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     This solution is now valid whenever Δ/2≦x≦1−Δ/2. Because the value of x is unknown, the sign of CE−CL is checked. If the sign is positive, Equation 7 is used. If the sign is negative, Equation 4 is used. This is written as follows: 
                       x   5     ⁡     (       C   ⁢           ⁢   E     ,     C   ⁢           ⁢   L     ,   Δ     )       =       1   -       Δ   /   2     ⁢         C   ⁢           ⁢   E     +     C   ⁢           ⁢   L           C   ⁢           ⁢   E     -     C   ⁢           ⁢   L         ⁢           ⁢   for   ⁢           ⁢     (       C   ⁢           ⁢   E     -     C   ⁢           ⁢   L       )         &gt;   0             (   8   )                   x   5     ⁡     (       C   ⁢           ⁢   E     ,     C   ⁢           ⁢   L     ,   Δ     )       =         -   1     -       Δ   /   2     ⁢         C   ⁢           ⁢   E     +     C   ⁢           ⁢   L           C   ⁢           ⁢   E     -     C   ⁢           ⁢   L         ⁢           ⁢   for   ⁢           ⁢     (       C   ⁢           ⁢   E     -     C   ⁢           ⁢   L       )         ≤   0             (   9   )               
with the subscript 5 indicating that we have combined the two solutions previously designated by subscripts 3 and 4.
 
     Equations 8 and 9 are plotted as shown in  FIG. 5 . 
     In this case, for the regions where the appropriate model is used, the desired linear behavior is achieved. In the central portion of the plot, where the correlator positions straddle the peak rather than both being on the same side of the peak, an undesirable answer is achieved. Equation 1 is always negative when x is to the left of the origin, and always positive when x is to the right of the origin. To the left of the origin, the more negative of Equations 1 and 9 is desired. To the right of the origin, the more positive of Equations 1 and 8 is desired. Therefore:
 
Discrim=Min[Eq. 1,Eq. 9] for ( CE−CL )≦0   (10)
 
Discrim=Max[Eq. 1,Eq. 8] for ( CE−CL )&gt;0   (11)
 
     As above, the result is plotted across the entire range from minus one code chip interval to plus one code chip interval (see  FIG. 6 ). The desired linear behavior across the range from −1+Δ/2 to 1−Δ/2 is attained, with saturation setting in outside of this range. 
       FIG. 7  shows two curves (both for the case where Δ=0.2 code chip intervals)—one curve is for the Equation 1 solution (diverges from linearity and provides an underestimate of the actual misalignment between the arriving C/A code and the locally generated C/A code outside of the central linear region) and the linear dotted line is for Equations 10 and 11 solutions (linear behavior over a range from −0.9 to 0.9). This is a nine-fold increase in the linear operating range. As a result, a code loop is able to converge more rapidly to the correct alignment of the code. 
       FIG. 8  shows curves similar to those shown in  FIG. 7  but when Δ is reduced to 0.1 code chip intervals. The amount of improvement provided by this technique increases as Δ decreases. 
     This linear result provides a direct relationship between the measurement and the amount of adjustment needed to make to the phase (temporal delay) of the local code generator. From the measurement, one can compute the amount of correction needed without being hindered by the departure from linearity that occurs in the conventional prior art. If the discriminator value comes out to be a value of x, then the local code generator needs to shift by that amount. Noise consideration may require that all corrections are reduced by some proportionality constant. But the key is, the present invention will always be closer to the correct adjustment value and hence will pull in faster. 
     While the preferred embodiment of the invention has been illustrated and described, as noted above, many changes can be made without departing from the spirit and scope of the invention. Accordingly, the scope of the invention is not limited by the disclosure of the preferred embodiment. Instead, the invention should be determined entirely by reference to the claims that follow.