Abstract:
An amplifier having an improved output current drive capability includes an input stage and an output stage. An input of the output stage is operatively coupled to an output of the input stage. The amplifier further includes a current regeneration circuit operatively coupled to the input of the output stage in a feedback arrangement, the current regeneration circuit feeding back a current to the output circuit in accordance with a predetermined scale factor, the fed back current being proportional to an input current supplied to the output stage. The input current supplied to the output stage is dynamically adjustable by the current regeneration circuit in response to an input current requirement at the output stage.

Description:
FIELD OF THE INVENTION  
         [0001]    The present invention relates generally to amplifiers, and more particularly relates to techniques for improving an output drive capability in an output stage of an amplifier without proportionally increasing an input current requirement of the output stage.  
         BACKGROUND OF THE INVENTION  
         [0002]    In certain amplifier or driver applications, such as, for example, in an asymmetric digital subscriber line (ADSL) system, the output current and voltage requirements may be too high for conventional complementary metal-oxide-semiconductor (CMOS) drivers. Consequently, for such applications, bipolar drivers are typically employed. However, in order to meet the speed and/or output drive specifications required by the ADSL system, traditional ADSL drivers designed using bipolar technology generally require an output stage bias current that is large enough to supply a worst case base current to the bipolar output stage transistors. This has traditionally been accomplished by designing the driver to have a quiescent bias current which is large enough to deliver the maximum load current anticipated, which, in a typical ADSL application, may be about 600 milliamperes (mA) or more at a frequency of at least 1 megahertz (MHz).  
           [0003]    If the current gain of the output transistors comprising the driver is large (e.g., 100), the input base current required by the driver will be relatively small. However, depending upon semiconductor fabrication process and/or temperature variations, for example, the current gain of the bipolar transistor devices may significantly decrease, and therefore the driver must be designed for such worst case current gain, thus causing the driver to dissipate an undesirable amount of quiescent current under normal operation. For instance, the worst case current gain of a typical npn transistor may be as low as about ten, while the worst case current gain of a pnp transistor can be as low as five. As the current gain of the driver output transistors decreases, the amount of input base current required to deliver the maximum load current must increase proportionally.  
           [0004]    Conventional techniques for reducing the quiescent current in an amplifier or driver have been proposed. These techniques, however, are generally not always sufficient or fully effective, and may not be practical, feasible, or otherwise cost-effective to implement in a given application. Accordingly, there exists a need for techniques for increasing an output drive capability of an amplifier without proportionally increasing the amount of input quiescent current required by an output stage of the amplifier.  
         SUMMARY OF THE INVENTION  
         [0005]    The present invention provides techniques for improving an output drive capability in an output stage of an amplifier without proportionally increasing an input quiescent current requirement of the output stage. By utilizing a current regeneration technique, the present invention essentially monitors an input bias current to the output stage and dynamically adjusts the input bias current in response to output load requirements.  
           [0006]    In accordance with one aspect of the invention, an amplifier having an improved output current drive capability includes an input stage and an output stage. An input of the output stage is operatively coupled to an output of the input stage. The amplifier further includes a current regeneration circuit operatively coupled to the input of the output stage in a feedback arrangement, the current regeneration circuit feeding back a current to the output circuit in accordance with a predetermined scale factor, the fed back current being proportional to an input current supplied to the output stage. The input current supplied to the output stage is dynamically adjustable by the current regeneration circuit in response to an input current requirement at the output stage.  
           [0007]    In accordance with another aspect of the invention, the amplifier includes a slew rate enhancement circuit operatively coupled to the input stage of the amplifier. The slew rate enhancement circuit preferably provides an alternate current path for charging and discharging a compensation circuit included in the amplifier for stabilizing the current regeneration circuit.  
           [0008]    These and other features and advantages of the present invention will become apparent from the following detailed description of illustrative embodiments thereof, which is to be read in connection with the accompanying drawings.  
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0009]    [0009]FIG. 1 is schematic diagram illustrating an exemplary amplifier including a regenerative biasing circuit, formed in accordance with the present invention.  
         [0010]    [0010]FIG. 2 is a graphical representation illustrating a relationship between output load current and input current in an amplifier in accordance with the invention. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0011]    The present invention will be described herein in the context of an illustrative amplifier circuit which may be used, for example, in an asynchronous digital subscriber line (ADSL) application. It should be appreciated, however, that the present invention is not limited to this or any particular amplifier circuit. Rather, the invention is more generally applicable to any suitable circuit in which it is desirable to improve an output drive capability and/or slew rate of the circuit without proportionally increasing an input bias current of the circuit. The term “amplifier” as used herein essentially refers to a circuit for multiplying an input signal applied to the circuit by a predetermined gain. Thus, an amplifier formed in accordance with the present invention may function as a buffer, for example, when the gain is set to one. Moreover, although implementations of the present invention are described herein using npn and pnp bipolar junction transistor (BJT) devices, it is to be appreciated that one or more of the transistors may be replaced by other suitable devices, such as, for example, composite npn-pnp devices or complementary metal-oxide-semiconductor (CMOS) devices, with or without modifications to the circuit, as understood by those skilled in the art.  
         [0012]    Since metal-oxide-semiconductor (MOS) transistors are voltage-controlled devices rather than current-controlled devices, the regenerative current biasing techniques of the present invention are more suitable for use with BJT devices, at least as used in an output stage of an amplifier circuit, which generally requires a predetermined amount of base current to produce a given output current from the amplifier. Moreover, BJT devices typically have a greater current drive capability at higher frequencies (e.g., around 1 megahertz (MHz)) as compared to, for example, power CMOS devices which generally have an inherently large gate capacitance. In an ADSL application, BJT devices have a further advantage in that deep sub-micron CMOS devices, which are often employed in high-speed applications, are not well-suited for operation at relatively high power supply voltages, e.g., ten volts.  
         [0013]    [0013]FIG. 1 is a schematic diagram illustrating an exemplary amplifier  100  employing slew rate enhanced regenerative biasing, in accordance with one aspect of the invention. The amplifier  100  includes a non-inverting or positive input INP, an inverting or negative input INN and an output OUT. Thus, the amplifier may be considered a differential input amplifier. For ease of explanation, the exemplary amplifier  100  maybe grouped according to functional sub-circuits or stages, including an input stage  102 , a slew rate enhancement circuit  108 , a source regeneration circuit  106 , a sink regeneration circuit  104 , and an output stage  110 . Each of these circuits or stages is described in further detail below. It is to be appreciated that certain of these circuits may be combined or incorporated into one or more other circuits or stages. For example, the source regeneration circuit  106  and the sink regeneration circuit  104  may be incorporated into a single current regeneration circuit or into the output stage  110 .  
         [0014]    The input stage  102  is preferably a differential amplifier, such as, for example, an operational amplifier, having a differential input which forms the inputs INP, INN of the amplifier  100 . Input stage  102  is operatively connected to a positive voltage supply VP and to a negative voltage supply VN, such as ground. The input stage  102  may also incorporate one or more gain stages for providing the amplifier  100  with a predetermined amount of gain. A more detailed discussion of input and gain stages that may be suitable for use with the present invention can be found, for example, in the text by Alan B. Grebene,  Bipolar and MOS Analog Integrated Circuit Design,  John Wiley &amp; Sons, pp. 215-246 (1984), which is incorporated herein by reference. Accordingly, a detailed description of the input stage  102  will not be presented herein. It is to be appreciated that the input stage  102  is not limited to a differential input stage. Moreover, in applications where no amplification is required (e.g., a buffer), the input stage  102  may be configured to provide a gain of one.  
         [0015]    The output stage  110  of amplifier  100  preferably includes a pair of bipolar pnp transistors B 100  and B 111 , and a pair of bipolar npn transistors B 101  and B 110 , each of the transistors having an emitter terminal (E), a base terminal (B) and a collector terminal (C). As will be understood by those skilled in the art, corresponding pairs of output transistors B 110 , B 100  and B 111 , B 101  may be formed as composite transistor devices. A composite transistor device essentially integrates a pnp transistor with an npn transistor, the composite device having a higher current gain β C , given as β n ·β p , where β n  and β p  are the respective current gains of the npn and pnp transistors comprising the composite transistor. Transistors B 100  and B 110  may be viewed as constituting a current source portion of the output stage  110  and transistors B 101  and B 111  may be viewed as constituting a current sink portion of the output stage. Output transistors B 110  and B 111  are preferably coupled in a complementary or quasi-complementary Class AB configuration, as described below.  
         [0016]    With regard to the current source portion of output stage  110 , transistors B 100 , B 110  are configured such that a current gain from the base terminal of transistor B 110  to the output OUT of the amplifier  100  is given as β B110 ·β B100 , where β B110  and β B100  are the current gains of transistors B 110  and B 100 , respectively. Specifically, the emitter terminal of transistor B 110  and the collector terminal of transistor B 100  are coupled to the output OUT of the amplifier, the emitter terminal of transistor B 100  is coupled to the positive voltage supply VP, and the base terminal of transistor B 100  is coupled to the collector terminal of transistor B 110  at node  122 .  
         [0017]    A bias circuit  126 , which may be implemented as a constant current source as shown, is connected between the positive voltage supply VP and node  122  and supplies a bias current  13  for biasing transistors B 100 , B 110  to a predetermined direct current (DC) quiescent operating point. The bias circuit  126  may be, for example, a simple resistor or it may be an active device, such as, but not limited to, a transistor coupled to an appropriate bias voltage source (not shown) in a conventional fashion. Furthermore, the bias circuit  126  may be configured to provide temperature-dependent biasing. This can be accomplished, for example, by including a bias voltage generator circuit (not shown) in amplifier  100  which produces a bias voltage which varies proportionally with temperature. In this manner, the amplifier may exhibit a substantially constant transconductance over a given temperature range.  
         [0018]    Similarly, with regard to the current sink portion of output stage  110 , transistors B 101 , B 111  are configured such that a current gain from the base terminal of transistor B 111  to the output OUT of the amplifier  100  is given by β B111 ·β B101 , where β B111  and β B101  are the current gains of transistors B 111  and B 101 , respectively. Specifically, the emitter terminal of transistor B 111  and the collector terminal of transistor B 101  are coupled to the output OUT of the amplifier, the emitter terminal of transistor B 101  is coupled to the negative voltage supply VN, and the base terminal of transistor B 101  is coupled to the collector terminal of transistor B 111  at node  124 . A bias circuit  128 , depicted as a conventional constant current sink, is connected between node  124  and the negative voltage supply VN and supplies a bias current  14  for biasing transistors B 101 , B 111  to a predetermined DC quiescent operating point. Bias circuit  124  may be formed in a manner consistent with bias circuit  126  previously described.  
         [0019]    With continued reference to FIG. 1, the base current for transistors B 110  and B 111  is preferably supplied, at least in part, by bias circuits  132  and  130 , respectively, which are depicted as conventional current sources. Bias circuit  132  is configured as a constant current source connected between the positive voltage supply VP and the base terminal of transistor B 110  at node  114  and supplies a bias current I 1  for biasing transistor B 110  to a predetermined DC quiescent operating point. Similarly, bias circuit  130  is configured as a constant current sink connected between the base terminal of transistor B 111  at node  116  and the negative voltage supply VN and supplies a bias current I 2  for biasing transistor B 111  to a predetermined DC quiescent operating point. The value of bias currents I 1  and I 2  will depend, at least in part, upon the quiescent operating point of the output stage transistors B 110 , B 111 .  
         [0020]    With specific regard to transistor B 110 , the base current I 1  supplied by bias circuit  132  is dynamically enhanced by source regeneration circuit  106  operatively coupled to the base terminal of transistor B 110 . The source regeneration circuit  106  preferably comprises an npn transistor B 124  and two pnp transistors B 126 , B 127 , each of the transistors having an emitter terminal (E), a base terminal (B) and a collector terminal (C). The emitter and base terminals of transistor B 124  are coupled to the emitter and base terminals, respectively, of output transistor B 110  such that transistor B 124  mirrors at least a portion of the collector current of transistor B 110  in accordance with a predetermined ratio. The emitter areas of transistors B 124  and B 110  are scaled by the predetermined ratio, such as, for example, 1:10 as shown in FIG. 1. The ratio is preferably chosen to substantially match a worst case current gain of corresponding transistor B 110 , which may be as low as about ten. By matching the ratio to the anticipated worst case current gain of transistor B 110 , transistor B 124  will produce a reference current through its collector terminal which closely approximates the worst case base current required by transistor B 110  to source a given output load current.  
         [0021]    The collector current of transistor B 124 , which, as previously stated, approximates the required worst case base current of transistor B 110 , is mirrored by transistors B 126 , B 127  which are configured as a conventional simple two-transistor current mirror. Specifically, transistor B 126  is connected in a diode arrangement, with the collector terminal of transistor B 126  coupled to its base terminal at node  118 . The collector terminal of transistor B 126  is coupled to the collector terminal of transistor B 124 , and the emitter terminal of transistor B 126  is connected to the positive voltage supply VP. Transistor B 127  is connected so that its emitter terminal is coupled to positive voltage supply VP and its base terminal is coupled to the base terminal of transistor B 126  at node  118 . The collector terminal of transistor B 127  is connected to the base terminal of transistor B 110 , such that the mirrored reference current in transistor B 124  is operatively fed back to the base terminal of transistor B 110 . Various conventional alternative current mirrors, such as, for example, a Wilson or Widlar current mirror, as understood by those skilled in the art, may also be employed by the present invention. A more detailed description of such current mirrors may be found, for example, in the text by Paul R. Gray and Robert G. Meyer,  Analysis and Design of Analog Integrated Circuits, Second Edition,  John Wiley &amp; Sons, pp. 233-246 (1984), which is incorporated herein by reference.  
         [0022]    Preferably, the emitter areas of transistors B 126  and B 127  are scaled to be substantially matched to one another (i.e., an emitter area ratio of 1:1). The present invention, however, contemplates that the emitter area ratio between transistors B 126  and B 127  may be chosen to be any number n (e.g., n:1), where n is greater than zero. In this instance, the emitter area ratio between transistors B 124  and B 110  should also be adjusted, such as, for example, 1:10n.  
         [0023]    By way of illustration, as the output load current sourced by the amplifier  100  increases, the collector current in transistor B 110  also increases which, in turn, increases the reference collector current in transistor B 124  by substantially the same proportion. Increasing the collector current in transistor B 124  increases the current in the mirror comprised of transistors B 126  and B 127 . The current in transistor B 127  is then fed back to the base terminal of transistor B 110 . Thus, the load current regenerates itself in the form of input base current such that the output stage  110  never starves due to a lack of base current drive. Moreover, a large quiescent current from bias circuit  132  is not required since the base current for transistor B 110  is generated dynamically by the source regeneration circuit  106 . Consequently, bias circuit  132  may be advantageously designed with a significantly smaller current than if source regeneration stage  106  were not present in the amplifier  100 .  
         [0024]    With continued reference to FIG. 1, the sink regeneration circuit  104  may be implemented in a manner consistent with the source regeneration circuit  106  described above. Regarding output transistor B 111 , the base current I 2  supplied by bias circuit  130  is dynamically enhanced by sink regeneration circuit  104  which is operatively coupled to the base terminal of transistor B 111 . The sink regeneration circuit  104  preferably comprises a pnp transistor B 125  and two npn transistors B 128 , B 129 , each of the transistors having an emitter terminal (E), a base terminal (B) and a collector terminal (C). The emitter and base terminals of transistor B 125  are coupled to the emitter and base terminals, respectively, of output transistor B 111  such that transistor B 125  mirrors at least a portion of the collector current of transistor B 111  in accordance with a predetermined ratio. The emitter areas of transistors B 125  and B 111  are scaled by the predetermined ratio, which, in the exemplary amplifier  100 , is selected to be 1:5.  
         [0025]    As in the case for transistors B 124  and B 110  described above, the emitter area ratio between transistors B 125  and B 111  is preferably chosen to substantially reflect an anticipated worst case current gain of output transistor B 111 . Since transistor B 111  is a pnp-type device, the worst case current gain may be as low as about five. By substantially matching the emitter area ratio to the anticipated worst case current gain of transistor B 111 , a reference collector current in transistor B 125  will closely approximate the base current required by transistor B 111  to supply a given output load current. It is to be appreciated that since the worst case current gain for a pnp transistor may be different from the worst case current gain for an npn transistor operating at a similar quiescent bias point, the emitter area ratio between the source regeneration circuit  106  and the source output transistor B 110  may not necessarily be the same as the emitter area ratio between the sink regeneration circuit  104  and the sink output transistor B 111 .  
         [0026]    The collector current of transistor B 125 , which, as previously stated, approximates the required base current of transistor B 111 , is mirrored by transistors B 128 , B 129  which are configured as a conventional simple current mirror. Specifically, transistor B 128  is connected in a diode arrangement, with the collector terminal of transistor B 128  coupled to its base terminal at node  120 . Various alternative current mirror arrangements are also contemplated by the present invention. The collector terminal of transistor B 128  is coupled to the collector terminal of transistor B 125 , and the emitter terminal of transistor B 128  is connected to the negative voltage supply VN. Transistor B 129  is coupled so that its emitter terminal is coupled to the negative voltage supply VN and its base terminal is coupled to the base terminal of transistor B 128  at node  120 . The collector terminal of transistor B 129  is connected to the base terminal of transistor B 111 , such that the mirrored reference collector current in transistor B 125  is fed back to the base terminal of transistor B 111  in a manner consistent with that previously explained in connection with transistor B 110 .  
         [0027]    Preferably, the emitter areas of transistors B 128  and B 129  are scaled to be substantially matched to one another (i.e., an emitter area ratio of 1:1), as in the case of transistors B 126 , B 127 . The present invention, however, contemplates that the emitter area ratio between transistors B 128  and B 129  maybe chosen to be any number n (e.g., n:1), where n is a predetermined number greater than zero. In this instance, the emitter area ratio between transistors B 125  and B 111  must also be adjusted, such as, for example, 1:5·n. It is to be appreciated that the emitter area ratio between transistors B 128 , B 129  is not dependent upon the emitter area ratio between transistors B 126 , B 127 .  
         [0028]    By way of illustration, as the output load current sunk by the amplifier  100  increases, the collector current in transistor B 111  also increases which, in turn, increases the collector current in transistor B 125  by the same proportion. Increasing the collector current in transistor B 125  similarly increases the current in the mirror comprised of transistors B 128 , B 129 . The current in transistor B 129  is fed back to the base terminal of transistor B 111 . Thus, the load current regenerates itself in the form of input base current such that the output stage  110  never starves due to a lack of base current. Consequently, bias circuit  130  maybe advantageously designed with a significantly smaller current than if sink regeneration circuit  104  were not present in the amplifier  100 .  
         [0029]    Referring again to FIG. 1, in order to stabilize the feedback loop in both the source regeneration circuit  106  and the sink regeneration circuit  104 , a compensation circuit comprised, for example, of capacitors C 1  and C 2 , is preferably included in amplifier  100 , capacitor C 1  being coupled between the positive voltage supply VP and the base terminal of transistor B 110  at node  114 , and capacitor C 2  being coupled between the base terminal of transistor B 111  at node  116  and the negative voltage supply VN. Without the use of such compensation circuit, the source and sink regeneration circuits may produce undesirable frequency peaking and/or oscillation at certain high frequencies. A preferred value for compensation capacitors C 1 , C 2  for use with the exemplary amplifier  100  is from about 10 picofarad (pF) to about 30 pF, and, more preferably, is about 17 pF. It is to be appreciated that compensation capacitors C 1 , C 2  do not have to be matched to one another, since each compensation capacitor C 1 , C 2  corresponds to a separate feedback loop to be compensated. In fact, compensation capacitors C 1  and C 2  maybe different in value due, at least in part, to differences in the small signal characteristics of the two regeneration circuits  104 ,  106 .  
         [0030]    Under certain conditions, particularly when large output signal swings at high frequency (e.g., about 1 MHz) are required, the output stage  110  may exhibit a nonlinear or unsymmetrical response, and thus the output signal may have higher distortion than is desirable. This nonlinearity may be attributed, at least in part, to a decreased slew rate as a result of the compensation circuit and/or capacitive load coupled to the output stage  110 . In order to increase the slew rate of the exemplary amplifier  100 , which is typically defined as the maximum output voltage rate, either positive or negative, a slew rate enhancement circuit  108  is preferably included in the amplifier. The slew rate of amplifier  100  is determined by the amount of current that can be sourced or sunk into an output/compensation capacitor. With respect to the compensation capacitors C 1 , C 2 , as the compensation capacitance increases in value, the slew rate of the amplifier will decrease proportionally. Consequently, the slew rate enhancement circuit  108  preferably provides an alternate current path for charging and discharging the compensation capacitors C 1 , C 2 . This alternate path preferably operates in a Class AB mode, that is, it comes into conduction primarily when large signal swings are involved. The present invention contemplates various other slew rate enhancement techniques.  
         [0031]    As shown in FIG. 1, the slew rate enhancement circuit  108  includes a pair of pnp transistors B 118 A, B 118 B and a pair of npn transistors B 119 A, B 119 B, each having an emitter terminal (E), a base terminal (B) and a collector terminal (C). The base terminals of each of the transistors B 118 A, B 118 B, B 119 A, B 119 B form an input of the slew rate enhancement circuit  108  which is coupled to an output of the input stage  102  at node  112 . The emitter terminals of transistors B 118 A and B 118 B are coupled together at node  114 . Similarly, the emitter terminals of transistors B 119 A and B 119 B are coupled together at node  116 . Each of the corresponding transistors B 118 A, B 118 B and B 119 A, B 119 B in a given pair of transistors are ideally matched, at least in terms of emitter area (i.e., an emitter area ratio of 1:1). The collector terminals of transistors B 118 A and B 119 A are coupled together to the output OUT of the amplifier, while the collector terminals of transistors B 118 B and B 119 B are connected in a cross-coupled arrangement to the sink regeneration stage and source regeneration stage at nodes  120  and  118 , respectively. Specifically, the collector terminal of transistor B 118 B is coupled to the base and collector terminals of diode-connected transistor B 128 , and the collector terminal of transistor B 119 B is coupled to the base and collector terminals of diode-connected transistor B 126 .  
         [0032]    By way of illustration, consider a positive voltage ramp presented at the input of the slew rate enhancement circuit  108  at node  112 . As the voltage at node  112  increases, the emitter voltage of transistors B 119 A, B 119 B will increase by essentially the same amount, since the base-emitter voltage of a bipolar transistor is relatively constant, despite small changes in current through the transistor. Assuming the voltage across compensation capacitor C 2  was initially zero, if the voltage at node  112  increases rapidly, the current produced by transistors B 119 A, B 119 B will increase significantly, thereby rapidly charging capacitor C 2 . Half of the charging current attributed to transistor B 119 B will be pulled from the current mirror comprising transistors B 126 , B 127  in source regeneration stage  106 , thereby rapidly discharging compensation capacitor C 1  by the collector current in transistor B 127 . The opposite holds true when considering a negative voltage ramp presented at node  112 . Accordingly, the slew rate enhancement circuit  108  increases the slew rate of the amplifier  100  without substantially increasing the quiescent bias current of the amplifier and without compromising the stability of the amplifier by requiring a smaller value compensation capacitor.  
         [0033]    With reference now to FIG. 2, a graphical representation is shown illustrating a relationship between the base current (IBB 110 ) of output transistor B 110  (y-axis) and the output current (ILOAD) for the exemplary amplifier  100  (x-axis). As is apparent from the figure, when the output current is 600 mA, the worst case input base current required for output transistor B 110  is about 1.5 mA, which occurs with low current gain (beta) transistors at low temperature (e.g., −40 degrees Celsius), as shown by curve  202 . Even at 25 degrees Celsius, the base current required to output 600 mA is about 0.5 mA, as shown by curve  204 . Without the source and sink regeneration stages of the present invention, the bias circuit  132  (see FIG. 1) must generate a current I 1  of at least 1.5 mA in order for the output stage to function properly during worst case conditions. Using the current regeneration techniques of the present invention described herein, the bias current I 1  can be as low as 100 microamperes (μA) and still produce the same output current of 600 mA, thus saving 1.4 mA of quiescent current. This is shown by curve  206  on the graph of FIG. 2.  
         [0034]    Although illustrative embodiments of the present invention have been described herein with reference to the accompanying drawings, it is to be understood that the invention is not limited to those precise embodiments, and that various other changes and modifications may be made therein by one skilled in the art without departing from the scope or spirit of the invention.