Abstract:
An RFID backscatter interrogator for transmitting data to an RFID tag, generating a carrier for the tag, and receiving data from the tag modulated onto the carrier, the interrogator including a single grounded-coplanar wave-guide circuit board and at least one surface mount integrated circuit supported by the circuit board.

Description:
TECHNICAL FIELD  
         [0001]    The invention relates to RFID readers. The invention also relates to grounded co-planar wave guide RFID readers. The invention also relates to portable RFID readers.  
         BACKGROUND OF THE INVENTION  
         [0002]    Remote communication utilizing wireless equipment typically relies on radio frequency (RF) technology, which is employed in many industries. One application of RF technology is in locating, identifying, and tracking objects, such as animals, inventory, and vehicles.  
           [0003]    RF identification (RFID) tag systems have been developed to identify, monitor, or control remote objects.  
           [0004]    An advantage of RFID systems is the non-contact, non-line-of-sight capability of the technology. Tags can be read through a variety of substances such as snow, fog, ice, paint, dirt, and other visually and environmentally challenging conditions where bar codes or other optically-read technologies would be useless. RF tags can also be read at remarkable speeds, in most cases responding in less than one hundred milliseconds.  
           [0005]    There are three main categories of RFID tag systems. These are systems that employ beam-powered passive tags, battery-powered semi-passive tags, and active tags. Each operates in fundamentally different ways. The invention described below in the Detailed Description can be embodied in any of these types of systems.  
           [0006]    The beam-powered RFID tag is often referred to as a passive device because it derives the energy needed for its operation from the radio frequency energy beamed at it. The tag rectifies the field and changes the reflective characteristics of the tag itself, creating a change in reflectivity (RF cross-section) that is seen at the interrogator. A battery-powered semi-passive RFID tag operates in a similar fashion, modulating its RF cross-section in order to change its reflectivity that is seen at the interrogator to develop a communication link. Here, the battery is the only source of the tag&#39;s operational power. Finally, in the active RFID tag, both the tag and reader have transceivers to communicate and are powered by respective batteries.  
           [0007]    A typical RF tag system will contain at least one tag and one interrogator. The range of communication for such tags varies according to the transmission power of the interrogator, interrogator receiver sensitivity and selectivity, and backscatter characteristics of the tag. Battery-powered tags operating at 2,450 MHz have traditionally been limited to less than ten meters in range. However, devices with sufficient power can reach in excess of 100 meters in range, depending on the frequency and environmental characteristics.  
           [0008]    Conventional continuous wave backscatter RF tag systems utilizing passive (no battery) RF tags require adequate power from a signal from the interrogator to power the internal circuitry in the tag used to modulate the signal back to the interrogator. While this is successful for tags that are located in close proximity to an interrogator, for example less than three meters, this may be insufficient range for some applications, for example greater than 100 meters.  
           [0009]    A coplanar waveguide is a transmission line that shares some characteristics with microstrip lines. The characteristic impedance of a coplanar waveguide transmission line is determined by the distributed inductance and the distributed capacitance from the strip to the adjacent groundplane. In a grounded coplanar waveguide, some of the fields go through air, and (ideally) only a small fraction leak to a groundplane. Because some of the fields are in air, there is less loss. Tuning of the dielectric while the circuit is on is possible with no risk of shorts. Large metal top surfaces improve heat sinking, and because the waveguide is grounded, metal and screws can be added for even more heatsinking. The coplanar waveguide can be used to mount components in series and to shunt without need for drilling or use of plated through holes. This makes some circuits possible which would not be possible using plated through holes, if the inductance of plated through holes to the groundplane would be too high. Frequency multipliers are easily used with coplanar waveguides because there is a topside ground to mount diodes in shunt.  
           [0010]    Readers or interrogators with good range have been developed by the assignee of the present invention using off-the-shelf packaged and connectorized components coupled together with coaxial interconnects. These readers have very long ranges, but are generally large, stationary, expensive units. A smaller, less expensive unit, with improved manufacturability, is desired.  
         SUMMARY OF THE INVENTION  
         [0011]    The invention provides an interrogator for transmitting data to an RFID tag, generating a carrier for the tag, and receiving data from the tag modulated onto the carrier, the interrogator comprising a single grounded-coplanar wave-guide circuit board and at least one surface mount integrated circuit supported by the circuit board.  
           [0012]    Another aspect of the invention provide an RFID backscatter interrogator for transmitting data to an RFID tag, generating a carrier for the tag, and receiving data from the tag modulated onto the carrier, the interrogator comprising a synthesizer integrated circuit including first and second RF synthesizers having respective synthesizer outputs; circuitry configured to toggle between the first and second RF synthesizers to effect frequency hopping; an AM/FM radio demodulator integrated circuit; an RF mixer, coupled between the synthesizer integrated circuit and the AM/FM radio demodulator integrated circuit, and configured to mix data with a carrier frequency which the FM demodulator integrated circuit is configured to demodulate; and a single grounded-coplanar wave-guide circuit board supporting at least the synthesizer integrated circuit, the AM/FM radio demodulator IC, and the RF mixer.  
           [0013]    Another aspect of the invention provides an RFID backscatter system comprising an interrogator for transmitting data to an RFID tag, generating a carrier for the tag, and receiving data from the tag modulated onto the carrier, the interrogator comprising a single grounded-coplanar wave-guide circuit board and at least one surface mount integrated circuit supported by the circuit board; and a tag in selective backscatter communication with the interrogator.  
           [0014]    Another aspect of the invention provides an RFID backscatter interrogator for transmitting data to an RFID tag, generating a carrier for the tag, and receiving data from the tag modulated onto the carrier, the interrogator comprising a transmitter including: a synthesizer integrated circuit including first and second RF synthesizers having respective synthesizer outputs; at least one RF switch coupled to the synthesizer integrated circuit; and at least one RF power amplifier coupled to the RF switch, wherein a transmitter is defined comprising the synthesizer integrated circuit, RF switch, and RF power amplifier, the power amplifier defining a local oscillator; and a receiver including: a low noise amplifier; circuitry coupled to the local oscillator defined by the power amplifier and configured to cancel a carrier from a signal received by the low noise amplifier; an AM/FM radio demodulator integrated circuit; and an RF mixer, coupled between the synthesizer integrated circuit and the AM/FM radio demodulator integrated circuit, and configured to mix data with a carrier frequency which the AM/FM demodulator integrated circuit is configured to demodulate.  
           [0015]    Another aspect of the invention provides an RFID backscatter interrogator for transmitting data to an RFID tag, generating a carrier for the tag, and receiving data from the tag modulated onto the carrier, the interrogator comprising a synthesizer integrated circuit including first and second RF synthesizers having respective synthesizer outputs; an AM/FM radio demodulator integrated circuit; and an RF mixer, coupled between the synthesizer integrated circuit and the AM/FM radio demodulator integrated circuit, and configured to mix data with a carrier frequency which the FM demodulator integrated circuit is configured to demodulate.  
           [0016]    Another aspect of the invention provides a dual band RFID backscatter interrogator for transmitting data to an RFID tag, generating a carrier for the tag, and receiving data from the tag modulated onto the carrier, the interrogator comprising a transmitter including: a synthesizer integrated circuit including first and second RF synthesizers having respective synthesizer outputs; a first section configured for transmission at a first frequency, including at least one RF switch switchably coupled to the synthesizer integrated circuit; and at least one RF power amplifier coupled to the RF switch, wherein a transmitter is selectively defined including the synthesizer integrated circuit, RF switch, and RF power amplifier; a second section configured for transmission at a second frequency different from the first frequency, including at least one RF switch switchably coupled to the synthesizer integrated circuit; and at least one RF power amplifier coupled to the RF switch of the second section, wherein a transmitter is selectively defined including the synthesizer integrated circuit, RF switch of the second section, and RF power amplifier of the second section; and a receiver including a third section configured for receiving backscatter at the first frequency, including a low noise amplifier; circuitry coupled to the at least one power amplifier of the first section and configured to cancel a carrier from a signal received by the low noise amplifier; an AM/FM radio demodulator integrated circuit; and an RF mixer, coupled between the synthesizer integrated circuit and the AM/FM radio demodulator integrated circuit, and configured to mix data with a carrier frequency which the FM demodulator integrated circuit is configured to demodulate; and a fourth section configured for receiving backscatter at the second frequency, including a low noise amplifier; circuitry coupled to the at least one power amplifier of the second section and configured to cancel a carrier from a signal received by the low noise amplifier of the fourth section; an AM/FM radio demodulator integrated circuit; and an RF mixer, coupled between the synthesizer integrated circuit and the AM/FM radio demodulator integrated circuit of the fourth section, and configured to mix data with a carrier frequency which the FM demodulator integrated circuit of the fourth section is configured to demodulate.  
           [0017]    Another aspect of the invention provides a dual band RFID backscatter interrogator for transmitting data to an RFID tag, generating a carrier for the tag, and receiving data from the tag modulated onto the carrier, the interrogator comprising a transmitter including: a synthesizer integrated circuit including first and second RF synthesizers having respective synthesizer outputs and used in operation for communication in a selected one of at least first and second RF bands; at least one RF switch coupled to the synthesizer integrated circuit; and at least one RF power amplifier coupled to the RF switch, wherein a transmitter is defined comprising the synthesizer integrated circuit, RF switch, and RF power amplifier; and a receiver including: a low noise amplifier; circuitry coupled to the at least one power amplifier and configured to cancel a carrier from a signal received by the low noise amplifier; an AM/FM radio demodulator integrated circuit; and an RF mixer, coupled between the synthesizer integrated circuit and the AM/FM radio demodulator integrated circuit, and configured to mix data with a carrier frequency which the FM demodulator integrated circuit is configured to demodulate; wherein the at least one RF switch, at least one RF power amplifier, low noise amplifier, circuitry configured to cancel a carrier, RF mixer, and AM/FM demodulator are broadbanded to cover the at least first and second bands. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0018]    Preferred embodiments of the invention are described below with reference to the following accompanying drawings.  
         [0019]    [0019]FIG. 1 is a block diagram of a conventional RFID communication system, including a tag and reader in which the invention could be incorporated.  
         [0020]    [0020]FIG. 2 is a circuit schematic of an RFID reader embodying various aspects of the invention.  
         [0021]    [0021]FIG. 3 is a circuit schematic of an RFID reader in accordance with one alternative embodiment.  
         [0022]    [0022]FIG. 4 is a circuit schematic of an RFID reader in accordance with another alternative embodiment of the invention. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0023]    Attention is directed to the following commonly assigned patent applications, which are incorporated herein by reference: U.S. Patent Application Attorney Docket 13094-B (BA4-094) entitled “Radio Frequency Identification Device Communications Systems, Wireless Communication Devices, Wireless Communication Systems, Backscatter Communication Methods, A Radio Frequency Identification Device and A Radio Frequency Identification Device Communication Method” by inventors Mike A. Hughes and Richard M. Pratt; U.S. Patent Application Attorney Docket 12961-B (BA4-095), entitled “Method of Simultaneously Reading Multiple Radio Frequency Tags, RF Tag, and RF Reader”, by inventors Emre Ertin, Richard M. Pratt, Mike A. Hughes, Kevin L. Priddy, and Wayne M. Lechelt; U.S. Patent Application Attorney Docket 13096-B (BA4-097), entitled “System and Method to Identify Multiple RFID Tags”, by inventors Mike A. Hughes and Richard M. Pratt; U.S. Patent Application Attorney Docket 13154-B (BA4-098), entitled “Wireless Communication Devices, Radio Frequency Identification Devices, Backscatter Communication Device Wake-Up Methods, Communication Device Wake-Up Methods and A Radio Frequency Identification Device Wake-Up Method”, by inventors Richard Pratt and Mike Hughes; U.S. Patent Application Attorney Docket 13218-B (BA4-099), entitled “Wireless Communication Systems, Radio Frequency Identification Devices, Methods of Enhancing a Communications Range of a Radio Frequency Identification Device, and Wireless Communication Methods”, by inventors Richard Pratt and Steven B. Thompson; U.S. Patent Application Attorney Docket 13219-B (BA4-100), entitled “Wireless Communications Devices, Methods of Processing a Wireless Communication Signal, Wireless Communication Synchronization Methods and a Radio Frequency Identification Device Communication Method”, by inventors Richard M. Pratt and Steven B. Thompson; U.S. Patent Application Attorney Docket 13252-B (BA4-101), entitled “Wireless Communications Systems, Radio Frequency Identification Devices, Wireless Communications Methods, and Radio Frequency Identification Device Communications Methods”, by inventors Richard Pratt and Steven B. Thompson; U.S. Patent Application Attorney Docket 13097-B (BA4-102), entitled “A Challenged-Based Tag Authentication Model, by inventors Mike A. Hughes” and Richard M. Pratt; U.S. patent application Ser. No. 09/589,001, filed Jun. 6, 2000, entitled “Remote Communication System and Method”, by inventors R. W. Gilbert, G. A. Anderson, K. D. Steele, and C. L. Carrender; U.S. patent application Ser. No. 09/802,408; filed Mar. 9, 2001, entitled “Multi-Level RF Identification System”; by inventors R. W. Gilbert, G. A. Anderson, and K. D. Steele; U.S. patent application Ser. No. 09/833,465, filed Apr. 11, 2001, entitled “System and Method for Controlling Remote Device”, by inventors C. L. Carrender, R. W. Gilbert, J. W. Scott, and D. Clark; U.S. patent application Ser. No. 09/588,997, filed Jun. 6, 2000, entitled “Phase Modulation in RF Tag”, by inventors R. W. Gilbert and C. L. Carrender; U.S. Patent application Ser. No. 09/589,000, filed Jun. 6, 2000, entitled “Multi-Frequency Communication System and Method”, by inventors R. W. Gilbert and C. L. Carrender; U.S. patent application Ser. No. 09/588,998; filed Jun. 6, 2000, entitled “Distance/Ranging by Determination of RF Phase Delta”, by inventor C. L. Carrender; U.S. patent application Ser. No. 09/797,539, filed Feb. 28, 2001, entitled “Antenna Matching Circuit”, by inventor C. L. Carrender; U.S. patent application Ser. No. 09/833,391, filed Apr. 11, 2001, entitled “Frequency Hopping RFID Reader”, by inventor C. L. Carrender.  
         [0024]    As shown in FIG. 1, an RFID system  10  includes an interrogator or reader  18  and transponders (commonly called RF tags)  16 . The interrogator  18  includes a transceiver with decoder  14  and one or more antennas, such as a transmit antenna  12  and receive antenna  13 . The tag  16  includes an antenna  24 . In operation, the transmit antenna  12  emits and the receive antenna  13  receives electromagnetic radio signals generated by the transceiver  14  to activate the tag  16 , and receive signals from the tag  16 . When the tag  16  is activated, data can be read from or written to the tag  16 .  
         [0025]    In some applications, the transceiver  14  and antenna  12  are components of an interrogator (or reader)  18 , which can be configured either as a hand-held or a fixed-mount device. The interrogator  18  emits the radio signals  20  in range from one inch to one hundred feet or more, depending upon its power output, the radio frequency used, and other radio frequency considerations. When an RF tag  16  passes through the electromagnetic radio waves  20 , the tag detects the signal  20  and is activated. Data encoded in the tag  16  is then transmitted by a modulated data signal  22  through the antenna  24  to the interrogator  18  for subsequent processing.  
         [0026]    The system  10  utilizes continuous wave backscatter to communicate data from the tag  16  to the interrogator  18 . More specifically, the interrogator  18  transmits a continuous-wave radio signal to the tag  16 , which modulates the signal  20  using modulated backscattering wherein the electrical characteristics of the antenna  24  are altered by a modulating signal from the tag that reflects a modulated signal  22  back to the interrogator  18 . The modulated signal  22  is encoded with information from the tag  16 . The interrogator  18  then demodulates the modulated signal  22  and decodes the information.  
         [0027]    [0027]FIG. 2 is a circuit schematic of an RF reader  30  embodying various aspects of the invention. The RF reader  30  of FIG. 2 could be used in place of the interrogator  18  of FIG. 1. In the illustrated embodiment, most major components of the RF reader  30  are included on a single printed circuit board  32 . More particularly, in the illustrated embodiment, the printed circuit board is a grounded co-planar wave guide (GCPWG) circuit board. This is a different technology than the technology that is typically used for a radio frequency printed circuit boards. Both micro-strip and grounded co-planar wave guide technologies have advantages over off the shelf, packaged technologies. They allow boards to be layed out as desired and tailored to a particular application. There are also large cost savings. Some packaged parts costing, for example, in the order of $200 or $500 can be replaced with surface mount parts costing in the order of $1.50 or $3. GCPWG technology allows for better performance and lower size and cost, even over micro-strip technology.  
         [0028]    The disclosed GCPWG design provides many advantages over traditional micro-strip techniques. GCPWG circuits are physically smaller in size for a given frequency, and offer greater noise immunity from on-board and off-board sources. GCPWG technology provides for easier transitions from surface mount devices to transmission lines on the printed circuit board while maintaining proper circuit impedance. With proper circuit impedance more closely maintained, transmission lines radiate less RF energy from the board, which makes the disclosed circuitry less susceptible to disturbance from the environment and less likely to cause any disturbance to adjacent circuitry.  
         [0029]    In the illustrated embodiment, the reader  30  operates in the ISM (Instrument, Scientific and Medical) band. The ISM Band is an unlicensed publicly owned part of the radio spectrum in the 900 MHz, 2.4 GHz and 5 GHz ranges. Variations of the reader  30  designed with GCPWG techniques can be advantageously used for any desired microwave ISM (Instrument, Scientific and Medical) range, e.g. 902 MHz to 928 MHz, 2.465 GHz to 2.4835 GHz and 5.785 GHz to 5.815 GHz. The ISM band provide advantages in that FCC (federal communication commission) licensing is not required. The reader  30  can be modified for use at frequencies outside the ISM band, if desired, for use in different countries or if licensing requirements are not a concern. Utilizing the disclosed GCPWG RFID reader at any frequency will allow for efficient RF board designs. GCPWG designs are more suited to smaller high performance stationary readers and high performance hand held readers than micro-strip designs or designs that utilize pre-packaged RF parts.  
         [0030]    The reader  30  includes a transmit section  34  and a receive section  36 . Some components could be described as being used by both a receive section or a transmit section, or arguably should be classified as belonging in a different section than shown. It should be kept in mind that the illustrated division between receive section  36  and transmit section  34  is a rough generalization. To better enable one of ordinary skill in the art to make and use the invention without undue experimentation, part numbers will be provided for one preferred embodiment of the invention. It should be kept in mind that alternative components from alternative manufacturers could be used. Additionally, functionality of one integrated circuit could be obtained by combining two or more integrated circuits or components; conversely, it is possible that functionality of two or more integrated circuits illustrated could be combined in an alternative single integrated circuit.  
         [0031]    The transmit section  34  includes a synthesizer  38 . The synthesizer  38  receives a reference signal from a reference generator  39  and performs frequency synthesis. In the illustrated embodiment, the synthesizer  38  is a single integrated circuit that includes two RF synthesizers that can be set independently of each other. In the illustrated embodiment, the synthesizer  38  is an integrated circuit designated Si4133 and sold by Silicon Laboratories, Inc., 4635 Boston Lane, Austin, Tex. 78735.  
         [0032]    The Si4133 is a monolithic integrated circuit that performs IF and dual-band RF frequency synthesis for wireless communications applications. The Si4133 includes three voltage-controlled oscillators, loop filters, reference and VCO dividers, and phase detectors. This reduces the need for external components. This integrated circuit is more typically used in applications such as cell phones and wireless LANs.  
         [0033]    In the illustrated embodiment, the output of the synthesizer is toggled for faster frequency hopping operations than with present readers. More particularly, as one synthesizer is being used to transmit, a new frequency is set up in the other synthesizer. Then a toggling between synthesizers takes place and a new frequency can be set up again. One synthesizer is used for transmitting while the other is set up for the next transmit frequency. Frequency hopping is used for FCC reasons, so that no one frequency is used for too long, so as to avoid causing interference, and to avoid jamming by other interfering devices. Frequency hopping generally means using different carrier frequencies at different times. The carrier will hop around within an ISM band so that an interfering signal will hopefully be avoided at some frequencies. Frequency hopping is particularly useful in the 2.45 GHz band because of the leakages from microwave ovens. In one embodiment, control circuitry  37  is provided to cause the synthesizer  38  to frequency hop. In the illustrated embodiment, the control circuitry  37  is located on a separate digital board.  
         [0034]    The transmit section  34  further includes a buffer amplifier  40  having an input coupled to an RF output of the synthesizer  38  and having an output. The buffer amplifier adds isolation. In the illustrated embodiment, the buffer amplifier  40  is an integrated circuit designated RF2046 and sold by RF Micro Devices, Inc., 7625 Thorndike Road, Greensboro, N.C. 27409.  
         [0035]    The transmit section  34  further includes one or more RF switches  42  having an input coupled to the output of the buffer amplifier and having an output. In the illustrated embodiment, two RF switches are provided that are turned on and off together. Two switches are provided for increased isolation. In the illustrated embodiment, each RF switch  42  is an integrated circuit designated HMC194MS8 and sold by Hittite Microwave Corporation, 12 Elizabeth Drive, Chelmsford, Mass. 01824.  
         [0036]    The transmit section  34  further includes a first power amplifier  44  having an input coupled to the output of the RF switches  42  and having an output. In the illustrated embodiment, the power amplifier  44  is an integrated circuit designated HMC414MS8G and sold by Hittite Microwave Corporation, 12 Elizabeth Drive, Chelmsford, Mass. 01824.  
         [0037]    The transmit section  34  further includes a directional RF coupler  46  having an input coupled to the output of the first power amplifier  44  and having an output defining a local oscillator. In the illustrated embodiment, the RF coupler  46  is an integrated circuit designated 1H1304- 20 , and sold under the brand name Xinger by Anaren Microwave, Inc., 6635 Kirkville Road, East Syracuse, N.Y.  13057 . Part of the output from the first power amplifier  44  is coupled off through the directional coupler  46 . Power level is reduced and a signal is provided to the receive section  36  for a first downconversion, which will be described below.  
         [0038]    The transmit section  34  may further include a second power amplifier  48 . If amplification up to a desired level can be obtained using the first amplifier  44 , the second amplifier  48  can be omitted. When included, the second power amplifier  48  receives the rest of the output from the first power amplifier, via the coupler  46 . The second power amplifier amplifies up for transmission at up to about 1 Watt, for conformance with FCC part  15  rules. Other output power levels could be used as desired. For example, different power levels may be appropriate in different countries. In the illustrated embodiment, the second power amplifier  48  is a high power linear amplifier integrated circuit designated RF2126 and sold by RF Micro Devices, Inc., 7625 Thorndike Road, Greensboro, N.C. 27409.  
         [0039]    The transmit section  34  further includes a transmit antenna  50  coupled to the second power amplifier  48 . While other antenna designs could be employed, in the illustrated embodiment the antenna  50  is a fractal array.  
         [0040]    The receive section  36  includes a receive antenna  52 . In the illustrated embodiment, the receive antenna  52  is substantially similar to the transmit antenna  50 ; however, in alternative embodiments, different antenna designs could be used for the transmit and receive antennas.  
         [0041]    The receive section  36  further includes a low noise amplifier  54  having an input coupled to the antenna  52  and having an output. The low noise amplifier  54  boosts the signal received by the receive antenna  52 . In the illustrated embodiment, the low noise amplifier  54  is an integrated circuit designated HMC287MS8 and sold by Hittite Microwave Corporation, 12 Elizabeth Drive, Chelmsford, Mass. 01824.  
         [0042]    The receive section  36  further includes a 0 degrees splitter  56  having an input coupled to the output of the low noise amplifier  54  and having two outputs. The splitter  56  produces two signals that are in phase.  
         [0043]    A 90 degrees splitter  58 , on the board  30 , has an input coupled to the coupler  46  and has two outputs, 90 degrees apart. The splitter  58  thus produces a phase shift for one of the outputs relative to the input.  
         [0044]    The receive section  36  further includes mixers  60  and  62  having respective inputs and outputs. The mixer  60  receives a signal from one of the outputs of the 90 degree splitter and a signal from one of the outputs of the 0 degrees splitter. The mixer  62  receives a signal from the other of the outputs of the 90 degree splitter and a signal from one of the outputs of the 0 degrees splitter. In the mixer  60  that gets the two original in-phase signals, the portion of the coupled off transmit signal (the reference signal) and the in-phase receive signal, the output of this mixer is still relatively in-phase with the original transmit signal. In the mixer  62  that gets the portion of the coupled off transmit signal that has been shifted in phase by 90 degrees and the other in phase receive signal, the output of this mixer  62  is 90 degrees off of the other. It either leads or lags the other by 90 degrees.  
         [0045]    At the first downconversion mixers  60  and  62 , the reflected carrier with the data modulation on it from the tag is mixed with the coupled off portion of the transmit signal. Because the two carriers are essentially the same, they mostly cancel, leaving just the modulation. There is also some ambient noise and sometimes some components of the carrier left over; however, because the output of the mixers (the data) is at a lower frequency and the rest of the circuit is tuned for this lower frequency and not 2.45 GHz, problems are avoided in having any of the 2.45 GHz get through to the rest of the circuit.  
         [0046]    In the illustrated embodiment, the mixers are respective integrated circuits designated MXA-2701-7 and sold by Premier Devices, Inc., 1860 Hartog Dr., San Jose, Calif. 95131.  
         [0047]    The receive section  36  further includes an op-amp  64  coupled to the output of the mixer  60 , and an op-amp  66  coupled to the output of the mixer  62 . In one embodiment, the op-amps  64  and  66  are defined by a dual op-amp integrated circuit designated AD8042AR and sold by Analog Devices, Inc., Three Technology Way, Norwood, Mass. 02062.  
         [0048]    The receive section  36  further includes mixers  68  and  70 , having respective inputs and outputs. The mixer  68  receives one input signal from the output of the op-amp  64 . The mixer  68  receives another input signal from a local oscillator (L.O.) output of the synthesizer  38 , via a buffer amp  72  on the board  32 , and a resistive divider defined by resistors  74 ,  76 , and  78  on the board  32 . The mixer  70  receives one input signal from the output of the op-amp  66 . The mixer  70  receives another input signal from a local oscillator (L.O.) output of the synthesizer  38 , via a buffer amp  72  on the board  32 , and the resistive divider defined by the resistors  74 ,  76 , and  78  on the board  32 . These mixers  68  and  70  are used for upconverting data signals from the op amps  64  and  66 , respectively, to a frequency that can be used by demodulators  82  and  84  that will be described below in more detail. In the illustrated embodiment, the mixers  68  and  70  are respective integrated circuits designated MXB-0503-3 and sold by Premier Devices, Inc., 1860 Hartog Dr., San Jose, Calif. 95131.  
         [0049]    The receive section  36  further includes a local oscillator  80 . In the illustrated embodiment, the local oscillator  80  is a 77.76 MHz local oscillator. More particularly, in the illustrated embodiment, the local oscillator  80  is a voltage controlled crystal oscillator integrated circuit designated VSLD55JC and sold by Connor-Winfield Corp., 2111 Comprehensive Dr., Aurora, Ill., 60505. According to the manufacturer, this integrated circuit is designed for phased lock loop applications requiring low jitter and tight stability.  
         [0050]    The receive section  36  further includes demodulators  82  and  84 . The demodulator  82  has an input coupled to the output of the mixer  68 , and an input coupled to the local oscillator  80  via a resistive divider defined by resistors  86 ,  88 , and  90 . The demodulator  84  has an input coupled to the output of the mixer  70 , and an input coupled to the local oscillator  80  via the resistive divider defined by resistors  86 ,  88 , and  90 . In the illustrated embodiment, the demodulators  82  and  84  are respective integrated circuits designated SA676 and sold by Philips Semiconductors, Eindhoven, Netherlands. The demodulators  82  and  84  generate first and second outputs  92  and  94  (commonly referred to as I and Q outputs; i.e., in-phase and quadrature outputs).  
         [0051]    The SA676 integrated circuit includes a mixer (its front end). This mixer is used, in the illustrated embodiment, as a down-converter. The mixer mixes an incoming data signal that is riding on a carrier frequency compatible with the frequency input requirements of the SA676 integrated circuit with the output of the local oscillator  80 , which in the illustrated embodiment is 77.76 MHz. The resulting down-conversion is a 455 KHz carrier with the desired data on it. This is then amplified within the SA676 integrated circuit and then data is detected through the use of a quadrature detector within the SA676 integrated circuit. The detected data is then pinned out of the SA676 integrated circuit as audio. This same signal is also run through a comparator within the SA676 integrated circuit and pinned out as a Received Signal Strength Indicator (RSSI)  96  or  98 . The Received Signal Strength Indicator (RSSI) output is used, in one embodiment, for dynamic gain control of one or more of: LNA Gain, Receive Gain, Transmit Gain or a combination. In one embodiment, the dynamic gain control is performed by the control circuitry  37 .  
         [0052]    There is 90 degrees of phase difference between the outputs  92  and  94  so that when one channel (I or Q) is in a quadrature null, the other channel is not and has good data on it. In one embodiment, these two outputs  92  and  94  are summed or X-OR&#39;ed so that a constant reliable data steam is realized. More particularly, in one embodiment, the two outputs are summed or X-OR&#39;ed external of the board  32 . In one embodiment, the two outputs are summed or X-OR&#39;ed by the control circuitry  37 .  
         [0053]    In one embodiment, the reader  30  is a portable, handheld reader, and further includes a battery  100  coupled to supply power to the various integrated circuits of the board  32 . In one embodiment, a housing  102  supports the battery and encloses at least a portion of the circuit board  32 . In alternative embodiments, the reader uses AC power (e.g., includes a transformer) and is portable or stationary.  
         [0054]    While a specific design has been disclosed for a 2450 MHz reader, all the above described integrated circuits are also available for the 915 MHz ISM band. An alternative embodiment provides a 915 MHz reader of similar size, function, and construction for utilization with tags that operate in the 915 MHz ISM band. Other alternative frequencies are also possible, though different specific circuitry would be employed and functions may have to be moved from the specific blocks disclosed to other or additional blocks.  
         [0055]    [0055]FIGS. 3 and 4 are circuit schematics of RFID readers in accordance with alternative embodiments. More particularly, FIGS. 3 and 4 illustrate dual-band GCPWG readers. A dual band reader can be constructed in a variety of ways.  
         [0056]    The RF reader  130  of FIG. 3 could be used in place of the interrogator  18  of FIG. 1. In the illustrated embodiment, most major components of the RF reader  130  are included on a single printed circuit board  132 . More particularly, in the illustrated embodiment, the printed circuit board is a grounded co-planar wave guide (GCPWG) circuit board, like the embodiment of FIG. 2.  
         [0057]    The reader  130  includes a transmit section  134  and a receive section  136 . It should be kept in mind that the illustrated division between receive section  136  and transmit section  134  is a rough generalization.  
         [0058]    The transmit section  134  includes a synthesizer  138 . The synthesizer  138  receives a reference signal from a reference generator  139  and performs frequency synthesis. In the illustrated embodiment, the synthesizer  138  is a single integrated circuit that includes two RF synthesizers that can be set independently of each other.  
         [0059]    The transmit section  134  further includes a buffer amplifier  140  having an input coupled to an RF output of the synthesizer  138  and having an output. The buffer amplifier adds isolation.  
         [0060]    The transmit section  134  further includes one or more RF switches  142  having an input coupled to the output of the buffer amplifier and having an output. In the illustrated embodiment, it is assumed that two RF switches (one switch  142  is shown) are provided that are turned on and off together. Two switches are provided for increased isolation.  
         [0061]    The transmit section  134  further includes a first power amplifier  144  having an input coupled to the output of the RF switches  142  and having an output.  
         [0062]    The transmit section  134  further includes a directional RF coupler  146  having an input coupled to the output of the first power amplifier  144  and having an output. Part of the output from the first power amplifier  144  is coupled off through the directional coupler  146 . Power level is reduced and a signal is provided to the receive section  136  for a first downconversion, which will be described below.  
         [0063]    The transmit section  134  may further include a second power amplifier  148 . If amplification up to a desired level can be obtained using the first amplifier  144 , the second amplifier  148  can be omitted. If included, the second power amplifier  148  receives the rest of the output from the first power amplifier, via the coupler  146 . The second power amplifier provides increased power for transmission.  
         [0064]    The transmit section  134  further includes a transmit antenna  150  coupled to the second power amplifier  148 .  
         [0065]    The receive section  136  includes a receive antenna  152 . Broad band, dual band, or individual band antennas are used in alternative designs, each having their own merits. Antennas can be selectively physically switched out by a user, as desired, in one embodiment.  
         [0066]    The receive section  136  further includes a low noise amplifier  154  having an input coupled to the antenna  152  and having an output. The low noise amplifier  154  boosts the signal received by the receive antenna  152 .  
         [0067]    The receive section  136  further includes a 0 degrees splitter  156  having an input coupled to the output of the low noise amplifier  154  and having two outputs. The splitter  156  produces two signals that are in phase.  
         [0068]    A 90 degrees splitter  158 , on the board  130 , has an input coupled to the coupler  146  and has two outputs, 90 degrees apart. The splitter  158  thus produces a phase shift for one of the outputs relative to the input.  
         [0069]    The receive section  136  further includes mixers  160  and  162  having respective inputs and outputs. The mixer  160  receives a signal from one of the outputs of the 90 degree splitter and a signal from one of the outputs of the 0 degrees splitter. The mixer  162  receives a signal from the other of the outputs of the 90 degree splitter and a signal from one of the outputs of the 0 degrees splitter. In the mixer  60  that gets the two original in-phase signals, the portion of the coupled off transmit signal (the reference signal) and the in-phase receive signal, the output of this mixer is still relatively in-phase with the original transmit signal. In the mixer  162  that gets the portion of the coupled off transmit signal that has been shifted in phase by 90 degrees and the other in phase receive signal, the output of this mixer  162  is 90 degrees off of the other. It either leads or lags the other by 90 degrees.  
         [0070]    At the first downconversion mixers  160  and  162 , the reflected carrier with the data modulation on it from the tag is mixed with the coupled off portion of the transmit signal. Because the two carriers are essentially the same, they mostly cancel, leaving just the modulation.  
         [0071]    The receive section  136  further includes an op-amp  164  coupled to the output of the mixer  160 , and an op-amp  166  coupled to the output of the mixer  162 .  
         [0072]    The receive section  136  further includes mixers  168  and  170 , having respective inputs and outputs. The mixer  168  receives one input signal from the output of the op-amp  164 . The mixer  168  receives another input signal from a local oscillator (L.O.) output of the synthesizer  138 , via a buffer amp  172  on the board  132 , and, e.g., a resistive divider defined by resistors  174 , 176 , and  178  on the board  132 . The mixer  170  receives one input signal from the output of the op-amp  166 . The mixer  170  receives another input signal from a local oscillator (L.O.) output of the synthesizer  138 , via a buffer amp  172  on the board  132 , and, e.g., the resistive divider defined by the resistors  174 , 176 , and  178  on the board  132 . These mixers  168  and  170  are used for upconverting data signals from the op amps  164  and  166 , respectively, to a frequency that can be used by demodulators  182  and  184  that will be described below in more detail.  
         [0073]    The receive section  136  further includes a local oscillator  180 . In the illustrated embodiment, the local oscillator  180  is a voltage controlled crystal oscillator.  
         [0074]    The receive section  136  further includes demodulators  182  and  184 . The demodulator  182  has an input coupled to the output of the mixer  168 , and an input coupled to the local oscillator  180  via, e.g., a resistive divider defined by resistors  186 ,  188 , and  190 . The demodulator  184  has an input coupled to the output of the mixer  170 , and an input coupled to the local oscillator  180  via, e.g., the resistive divider defined by resistors  186 ,  188 , and  190 . In the illustrated embodiment, the demodulators  182  and  184  generate first and second outputs (commonly referred to as I and Q outputs; i.e., in-phase and quadrature outputs). The demodulators  182  and  184  include mixers that are used, in the illustrated embodiment, as down-converters. The demodulators  182  and  184  further produce a Received Signal Strength Indicator (RSSI)  196  or  198 . The Received Signal Strength Indicator (RSSI) output is used, in one embodiment, for dynamic gain control of one or more of: LNA Gain, Receive Gain, Transmit Gain or a combination. In one embodiment, the dynamic gain control is performed by the control circuitry  137 .  
         [0075]    There is 90 degrees of phase difference between the outputs from the demodulators  182  and  184  so that when one channel (I or Q) is in a quadrature null, the other channel is not and has good data on it. In one embodiment, these two outputs are summed or X-OR&#39;ed so that a constant reliable data steam is realized. In one embodiment, the two outputs are summed or X-OR&#39;ed by the control circuitry  137 .  
         [0076]    In the embodiment shown in FIG. 3, the front end RF components are broadbanded to cover both bands and the synthesizer  138  is a dual band synthesizer. In the embodiment shown in FIG. 3, the synthesizer has an output in the 2450 MHz band, an output in the 915 band, and a local oscillator output.  
         [0077]    The reader  130  is a portable, handheld reader, and further includes a battery  200  coupled to supply power to components of the board  132 . In one embodiment, a housing  202  supports the battery and encloses at least a portion of the circuit board  132 . In alternative embodiments, the reader uses AC power (e.g., includes a transformer) and is portable or stationary.  
         [0078]    In the embodiment of FIG. 3, maximum performance in each individual band is not realized due to the broadband design, but the performance may be acceptable for applications that do not require maximum performance but do require a reader that can read tags that operate in two different bands; e.g., 2450 MHz and 915 MHz. A reader that is compact and full featured is realized.  
         [0079]    An alternative dual band embodiment, shown in FIG. 4, also uses a dual band synthesizer, and has different front ends for different bands, and involves actively switching between the different front ends as required by the application.  
         [0080]    The RF reader  230  of FIG. 4 could be used in place of the interrogator  18  of FIG. 1. In the illustrated embodiment, most major components of the RF reader  230  are included on a single printed circuit board  232 . More particularly, in the illustrated embodiment, the printed circuit board is a grounded co-planar wave guide (GCPWG) circuit board, like the embodiment of FIG. 2.  
         [0081]    The reader  230  includes a first frequency section (e.g., 2.45 GHz), a second frequency section (e.g., 915 MHz), and a common section  228 .  
         [0082]    The common section  228  includes a synthesizer  238 . The synthesizer  238  receives a reference signal from a reference generator  239  and performs frequency synthesis. In the illustrated embodiment, the synthesizer  238  is a single integrated circuit that includes two RF synthesizers that can be set independently of each other. More particularly, in the embodiment shown in FIG. 4, the synthesizer has an output in the 2450 MHz band, an output in the 915 band, that are selectively switched, and a local oscillator output.  
         [0083]    The sections  226  and  227  include buffer amplifiers  240  and  241 , respectively, having respective inputs selectively coupled to an RF output of the synthesizer  238  and having respective outputs. More particularly, the reader  230  further includes band select switches  229  and  231 , on the board  232 , for selecting between the first and second bands. In the illustrated embodiment, the switches  229  and  231  are commonly controlled by a mechanical or electrical switch (not shown). The buffer amps  240  and  241  are coupled to the synthesizer  238  via the switch  229  for selection of one band or the other for transmission. The buffer amplifiers  240  and  241  add isolation.  
         [0084]    The sections  226  and  227  further include RF switches  242  and  243 , respectively. The RF switch  242  has an input coupled to the output of the buffer amplifier  240  and the RF switch  243  has an input coupled to the output of the buffer amplifier  241 . The RF switches  242  and  243  have respective outputs.  
         [0085]    The sections  226  and  227  further include power amplifiers  244  and  245 , respectively. The power amplifier  244  has an input coupled to the output of the RF switch  242  and has an output. The power amplifier  245  has an input coupled to the output of the RF switch  243  and has an output.  
         [0086]    The sections  226  and  227  further include directional RF couplers  246  and  247 , respectively. The coupler  246  has an input coupled to the output of the power amplifier  244  and has an output. The coupler  247  has an input coupled to the output of the power amplifier  245  and has an output. Part of the output from the power amplifiers  244  and  245  is coupled off through the directional couplers  246  and  247 . Power level is reduced and a signal is provided to respective receive sections for a first downconversion, which will be described below.  
         [0087]    The section  226  may further include a second power amplifier  248 . If amplification up to a desired level can be obtained using the amplifier  244 , the second amplifier  248  can be omitted. The second power amplifier  248  receives the rest of the output from the power amplifier  244 , via the coupler  246 . The second power amplifier  248  increases power for transmission.  
         [0088]    The section  226  further includes a transmit antenna  250  coupled to the second power amplifier  248 . The section  227  includes a transmit antenna  251  coupled to the output of the power amplifier  245 .  
         [0089]    The section  226  includes a receive section  236  and the section  227  includes a receive section  237 . Receive sections  236  and  237  include receive antennas  252  and  253 , respectively. Individual band antennas are used in the illustrated embodiment.  
         [0090]    The receive sections  236  and  237  further include low noise amplifiers  254  and  255 , respectively. The amplifier  254  has an input coupled to the antenna  252  and has an output. The amplifier  255  has an input coupled to the antenna  253  and has an output. The low noise amplifiers  254  and  255  boost the signals received by the receive antennas  252  and  253 , respectively.  
         [0091]    The receive sections  236  and  237  further include 0 degrees splitters  256  and  257 , respectively. The splitter  256  has an input coupled to the output of the low noise amplifier  254  and has two outputs. The splitter  257  has an input coupled to the output of the low noise amplifier  255  and has two outputs. The splitters  256  and  257  respectively produce two signals that are in phase.  
         [0092]    A 90 degrees splitter  258 , on the board  230 , has an input coupled to the coupler  246  and has two outputs, 90 degrees apart. The splitter  258  thus produces a phase shift for one of the outputs relative to the input. A 90 degrees splitter  259 , on the board  230 , has an input coupled to the coupler  247  and has two outputs, 90 degrees apart.  
         [0093]    The receive section  236  further includes mixers  260  and  262  having respective inputs and outputs. The mixer  260  receives a signal from one of the outputs of the 90 degree splitter  258  and a signal from one of the outputs of the 0 degrees splitter  256 . The mixer  262  receives a signal from the other of the outputs of the 90 degree splitter  258  and a signal from one of the outputs of the 0 degrees splitter  256 . In the mixer  260  that gets the two original in-phase signals, the portion of the coupled off transmit signal (the reference signal) and the in-phase receive signal, the output of this mixer is still relatively in-phase with the original transmit signal. In the mixer  262  that gets the portion of the coupled off transmit signal that has been shifted in phase by 90 degrees and the other in phase receive signal, the output of this mixer  262  is 90 degrees off of the other. It either leads or lags the other by 90 degrees.  
         [0094]    At the first downconversion mixers  260  and  262 , the reflected carrier with the data modulation on it from the tag is mixed with the coupled off portion of the transmit signal. Because the two carriers are essentially the same, they mostly cancel, leaving just the modulation.  
         [0095]    Similarly, the receive section  236  further includes mixers  261  and  263  having respective inputs and outputs. The mixer  261  receives a signal from one of the outputs of the 90 degree splitter  259  and a signal from one of the outputs of the 0 degrees splitter  257 . The mixer  262  receives a signal from the other of the outputs of the 90 degree splitter  259  and a signal from one of the outputs of the 0 degrees splitter  257 .  
         [0096]    The section  236  further includes an op-amp  264  selectively coupled to the output of the mixer  260  or  261 , via the switch  231 . The section  237  further includes an op-amp  266  selectively coupled to the output of the mixer  262  or  263 , via the switch  231 .  
         [0097]    The sections  236  and  237  respectively further include mixers  268  and  270 , having respective inputs and outputs. The mixer  268  receives one input signal from the output of the op-amp  264 . The mixer  268  receives another input signal from a local oscillator (L.O.) output of the synthesizer  238  via a resistive divider defined by resistors  274 ,  276 , and  278  on the board  232 . The mixer  270  receives one input signal from the output of the op-amp  266 . The mixer  270  receives another input signal from a local oscillator (L.O.) output of the synthesizer  238 , via the resistive divider defined by the resistors  274 ,  276 , and  278  on the board  232 . These mixers  268  and  270  are used for upconverting data signals from the op amps  264  and  266 , respectively, to a frequency that can be used by demodulators  282  and  284  that will be described below in more detail.  
         [0098]    The section  228  further includes a local oscillator  280 . In the illustrated embodiment, the local oscillator  280  is a voltage controlled crystal oscillator.  
         [0099]    The sections  228  and  237  respectively further include demodulators  282  and  284 . The demodulator  282  has an input coupled to the output of the mixer  268 , and an input coupled to the local oscillator  280  via, e.g., a resistive divider defined by resistors  286 ,  288 , and  290 . The demodulator  284  has an input coupled to the output of the mixer  270 , and an input coupled to the local oscillator  280  via, e.g., the resistive divider defined by resistors  286 ,  288 , and  290 . In the illustrated embodiment, the demodulators  282  and  284  generate first and second outputs (commonly referred to as I and Q outputs; i.e., in-phase and quadrature outputs). The demodulators  282  and  284  include mixers that are used, in the illustrated embodiment, as down-converters. The demodulators  282  and  284  further produce Received Signal Strength Indicators (RSSI)  296  and  298 . The Received Signal Strength Indicator (RSSI) outputs are used, in one embodiment, for dynamic gain control of one or more of: LNA Gain, Receive Gain, Transmit Gain or a combination. In one embodiment, the dynamic gain control is performed by control circuitry  235 .  
         [0100]    There is 90 degrees of phase difference between the outputs from the demodulators  282  and  284  so that when one channel (I or Q) is in a quadrature null, the other channel is not and has good data on it. In one embodiment, these two outputs are summed or X-OR&#39;ed so that a constant reliable data steam is realized. In one embodiment, the two outputs are summed or X-OR&#39;ed by the control circuitry.  
         [0101]    The reader  230  is a portable, handheld reader, and further includes a battery  300  coupled to supply power to components of the board  232 . In one embodiment, a housing  302  supports the battery and encloses at least a portion of the circuit board  232 . In alternative embodiments, the reader uses AC power (e.g., includes a transformer) and is portable or stationary.  
         [0102]    The embodiment shown in FIG. 4 involves a physically larger reader which would have more than double the front end components, but yields higher performance in each band. Again, this embodiment uses broadband antennas, dual band antennas, or individual band antennas that are physically switched out as required by specific applications.  
         [0103]    The disclosed designs provide a different approach for RF reader design that allows for excellent performance and sensitivity, small physical size, and low manufacturing costs. One embodiment of the invention provides a design that uses commercially available surface mount parts on a grounded co-planar wave-guide printed circuit board. Previously, various parts were separately packaged and physically large and expensive. These parts and the semi-rigid coaxial method of interconnect made for an even larger physical size of the reader and labor intensive assembly. The circuit design disclosed here advantageously selects and uses commercially available surface mount parts to reduce the size of coaxial interconnects and to keep several related RF parts and processes on one printed circuit board that is about 1/60th the size of a previous reader&#39;s RF section. By providing a GCPWG design for an RFID reader, a physically smaller printed circuit board is possible compared to micro-strip type printed circuit boards. GCPWG circuits also generate less stray radiated signals and are less susceptible to them.  
         [0104]    In compliance with the statute, the invention has been described in language more or less specific as to structural and methodical features. It is to be understood, however, that the invention is not limited to the specific features shown and described, since the means herein disclosed comprise preferred forms of putting the invention into effect. The invention is, therefore, claimed in any of its forms or modifications within the proper scope of the appended claims appropriately interpreted in accordance with the doctrine of equivalents.