Abstract:
A floating symmetrical current limiter device blocks large bipolar input signals to the input circuit of an instrumentation device by transitioning between a low-impedance mode and a high-impedance mode. The current limiter device includes a signal path and a control path that are each coupled between an input terminal and an output terminal. The signal path has a low impedance that passes small differential signals across the limiter from the input terminal to the output terminal. The control path is responsive to large bipolar signals that appear across the limiter terminals by transitioning between a voltage divider and a constant-current source-based bias that controls the impedance of the signal path to become a large impedance, thereby blocking the large bipolar input signal from the output terminal.

Description:
BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to a current limiter circuit for protecting input circuits. More particularly, the present invention relates to a current limiter circuit for protecting input circuits from excessive over-voltage conditions and excessive input currents, while providing low distortion for small-signal input voltages. 
   2. Description of the Related Art 
   Input circuits appear in a wide variety of applications, including instrumentation devices such as Digital Multimeters (DMMs), oscilloscopes, spectrum analyzers and general purpose data acquisition equipment. Typically, input protection is required for preventing input circuits from destruction caused by over-voltage conditions. 
   In many cases, a simple arrangement of diode clamps are utilized for limiting the input voltage to the internal power supplies of the circuit. Such an arrangement, however, creates a condition in which excessive current may be injected externally through the clamp diodes. 
     FIG. 1  shows a schematic block diagram of a typical input circuit  100  having a conventional current limiter  101  for limiting excessive current. Input circuit  100  includes an input resistor R 1 , a current limiter device  101 , two clamp diodes D 1  and D 2 , and an amplifier A 1 . An input signal input at IN is applied to resistor R 1 . The input signal is coupled through current limiter device  101  to the input of amplifier A 1 . Current limiter device  101  is depicted in  FIG. 1  as a resistor. The anode of clamp diode D 1  is coupled to the input of amplifier A 1 . The cathode of clamp diode D 1  is coupled to supply voltage V CC . The cathode of clamp diode D 2  is coupled to the input of amplifier A 1 . The anode of clamp diode D 2  is coupled to supply voltage V EE . Clamp diodes D 1  and D 2  limit the input voltage that can be applied to the input to amplifier A 1  to about supply voltages V CC  and V EE . Current input device  101  limits that amount of current that can be supplied externally to clamp diodes D 1  and D 2  and to the input of amplifier A 1  when the externally applied input voltage exceeds the clamping voltages of V CC  and V EE . 
     FIGS. 2A–2C  depict circuit components that are conventionally used as current limiting devices. For example,  FIG. 2A  depicts a resistor  201 .  FIG. 2B  depicts a Positive Temperature Coefficient (PTC) thermistor  202 .  FIG. 2C  depicts a light bulb  203 . 
   Another example of a conventional input protection circuit is disclosed by U.S. Pat. No. 5,742,463 to Harris. According to Harris, such an input protection circuit includes at least two depletion-mode field effect transistors, and can provide unipolar or bipolar operation, thereby protecting an input circuit from both positive-going and negative-going voltage transients. 
   The goals of an ideal current limiter include the capability to prevent destruction of input components, including the limiter itself. Input current for such an ideal current limiter should be limited based on the maximum expected input voltage. The ideal current limiter should also provide a low-noise, highly-linear, low-value impedance for normal, small-voltage operating conditions, while providing a high impedance for large voltages. Thus, the impedance state of an ideal current limiter must change based on the applied voltage. Additionally, both the inrush transient current and static power dissipation of an ideal current limiter should be minimized for preventing failure of any components. 
   What is needed is yet a better technique for limiting overload current for preventing destruction of clamp diodes and input circuitry. 
   BRIEF SUMMARY OF THE INVENTION 
   The present invention provides a current limiter circuit for limiting overload current and thereby preventing destruction of the input components of an input circuit, including the current limiter circuit itself The current limiter of the present invention is characterized by three regions of operation and provides a low-noise, highly-linear, low-value input impedance for normal, small-voltage operating conditions, while providing a protective high impedance for large voltages. The current limiter circuit is symmetrical and floats on the input signal without connection to ground or power supplies. Further, the inrush transient current and the static power dissipation of a current limiter according to the present invention are minimized. 
   The advantages of the present invention are provided by a current limiter device that has a signal path and a control path that are both coupled between an input terminal and an output terminal. The output terminal can be coupled to an input circuit of, for example, an instrumentation device, a digital multimeter, an oscilloscope, a spectrum analyzer or a general-purpose data-acquisition device. According to the invention, the signal path of the current limiter device has a low impedance that passes small differential signals from the input terminal to the output terminal for voltages that are typically less than about one volt across the limiter. The control path is responsive to larger bipolar signals applied across the limiter by outputting a substantially constant current that is considerably less than what would be present in the low impedance path. The substantially constant current controls the impedance of the signal path to be a large impedance, thereby blocking the large bipolar input signal from the output terminal. 
   An alternative embodiment of the present invention provides a current limiter circuit having a signal path and a control path that are each coupled between an input terminal and an output terminal. The output terminal can be coupled to an input circuit of, for example, an instrumentation device, a digital multimeter, an oscilloscope, a spectrum analyzer or a general-purpose data-acquisition device. The signal path includes at least one depletion-mode device, such as an N-channel depletion-mode MOSFET, and a variable-impedance device, such as a P-Channel JFET, and passes small differential signals from the input terminal to the output terminal for differential signals that are typically less than one volt. Additionally, the signal path has a low impedance for these small differential signals across the limiter. The control path includes at least one depletion-mode device, such as an N-channel depletion-mode MOSFET and outputs at least one substantially constant current in response to larger bipolar input signals applied across the limiter. Each substantially constant current is considerably less than would be present in the low impedance path and controls at least one depletion-mode device of the signal path to be a high-impedance device and the variable-impedance device to be a high-impedance device so that the large bipolar input signal is blocked from the output terminal. 
   Yet another alternative embodiment of the present invention provides a current limiter device having a first terminal, a second terminal, and a current limiter circuit that is coupled between the first terminal and the second terminal. The current limiter circuit has a substantially constant-resistance operating mode when the magnitude of a voltage differential between a voltage at the first terminal and a voltage at the second terminal is less than or equal to a first predetermined voltage differential. The current limiter circuit also has a substantially constant-current operating mode when the magnitude of the voltage differential between the voltage at the first terminal and the voltage at the second terminal is greater than or equal to a second predetermined voltage differential. Lastly, the current limiter circuit has a transition operating mode when the magnitude of the voltage differential between the voltage at the first terminal and the voltage at the second terminal is between the first and second predetermined voltage differentials. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention is illustrated by way of example and not by limitation in the accompanying figures in which like reference numerals indicate similar elements and in which: 
       FIG. 1  shows a schematic block diagram of a typical input circuit having a conventional current limiter; 
       FIGS. 2A–2C  depict circuit components that are conventionally used as current limiting devices; 
       FIG. 3  shows a schematic diagram of an exemplary embodiment of a current limiter circuit according to the present invention; 
       FIGS. 4A–4D  show equivalent circuit models for illustrating operation of the current limiter circuit shown in  FIG. 3  for small bipolar normal signals; 
       FIGS. 5A–5F  show equivalent circuit models for illustrating operation of the current limiter circuit shown in  FIG. 3  for large positive overload signals; 
       FIGS. 6A–6F  show equivalent circuit models for illustrating operation of the current limiter circuit shown in  FIG. 3  for large negative overload signals; 
       FIG. 7  shows an exemplary graph illustrating the three operating regions of the current limiter circuit shown in  FIG. 3 ; 
       FIG. 8  is a graph illustrating current as a function of voltage across the exemplary current limiter circuit shown in  FIG. 3  according to the present invention with respect to typical input characteristics for other conventional current limiting devices; 
       FIG. 9  is a graph illustrating power dissipation as a function of voltage across the exemplary current limiter circuit shown in  FIG. 3  according to the present invention with respect to typical input characteristics for other conventional current limiting devices; 
       FIG. 10  is a graph illustrating nonlinearity characteristics as a function of voltage across the exemplary current limiter circuit shown in  FIG. 3  with respect to a conventional input protection circuit; and 
       FIG. 11  is a graph illustrating inrush current characteristics as a function of a 100 Volt step across the exemplary current limiter circuit according to the present invention with respect to typical input characteristics for other conventional current limiting devices. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   The present invention provides a current limiting circuit that protects input circuits from excessive current. One exemplary embodiment of a current limiting circuit of the present invention provides a bipolar floating limiter having four depletion-mode N-channel MOSFET transistors. The bipolar floating limiter is characterized by three regions of operation and provides a linear low-impedance input for normal-level small signals and a constant current source for overload signals. The four depletion mode N-channel MOSFET transistors provide high voltage overload capability. A single P-Channel JFET provides foldback current limiting during overload conditions, thereby providing low power dissipation. Four resistors are used for configuring the limiter characteristics. Under normal small-signal operation, the current limiter circuit of the present invention is inherently linear because only resistors and FETs are used. 
     FIG. 3  shows a schematic diagram of an exemplary embodiment of a current limiter circuit  300  according to the present invention. Current limiter circuit  300  includes four depletion-mode N-channel MOSFET transistors Q 1 –Q 4 , a P-channel JFET transistor Q 5  and four resistors R 1 –R 4 , which together form two circuit paths. The first circuit path is formed by input terminal T 1  being coupled to the drain of transistor Q 1 . The substrate of transistor Q 1  is connected to the source of transistor Q 1 , and the source of transistor Q 1  is coupled to the drain of transistor Q 5 . The gate of transistor Q 1  is coupled to the source of transistor Q 5 , and to the source and substrate of transistor Q 2 . The gate of transistor Q 2  is coupled to the drain of Q 5 , and to the source and substrate of transistor Q 1 . The drain of transistor Q 2  is coupled to output terminal T 2 . Output terminal T 2  is typically coupled to an input circuit of an instrumentation device, such as a Digital Multimeter (DMM), an oscilloscope, a spectrum analyzer or a general-purpose data-acquisition equipment. 
   The second circuit path, a control path, is formed by input terminal T 1  being coupled to the drain of transistor Q 3 . The substrate of transistor Q 3  is connected to the source of transistor Q 3  and to one terminal of resistor R 3 . The gate of transistor Q 3  is coupled to the other terminal of resistor R 3  and to one terminal of resistor R 1 . The other terminal of resistor R 1  is coupled to the gate of transistor Q 5  and to one terminal of resistor R 2 . The other terminal of resistor R 2  is coupled to the gate of transistor Q 4  and to one terminal of resistor R 4 . The other terminal of resistor R 4  is coupled to the source and the substrate of transistor Q 4 . The drain of transistor Q 4  is coupled to output terminal T 2 . 
   Current limiter circuit  300  provides current limiting in a floating symmetrical bipolar fashion. Consequently, small signal operation of current limiter circuit  300  can be described by reference to the equivalent circuit models shown in  FIGS. 4A–4D .  FIG. 4A  shows a schematic diagram for current limiter circuit  400  for small bipolar limiter voltage V L  conditions, such that |V L |&lt;&lt;1 V. Under normal small-signal conditions, there is insufficient voltage between terminals T 1  and T 2  for producing the VgsOff voltage of transistors Q 3  and Q 4 . As such, resistors R 1 –R 4  hold the gate of transistor Q 5  near the mid-voltage of the terminal potentials, and the R ds  value of transistor Q 5  is simply its full conduction R ds  value. Under normal small-signal condition between terminals T 1  and T 2 , the R ds  of transistors Q 1  and Q 2  are also at full conduction.  FIG. 4B  shows a schematic diagram for an equivalent circuit model  401  showing that for |V L |&lt;&lt;1 V, all circuit components can be represented by resistances. Resistance values for resistors R 1 –R 4  are each typically greater than 10 kΩ, while the R ds  values for each transistor is typically less than 100 Ω. Thus, the normal-state resistance between terminals T 1  and T 2  for |V L |&lt;&lt;1 V is approximately R ds (Q 1 )+R ds (Q 5 )+R ds (Q 2 ), as represented by equivalent circuit  402  in  FIG. 4C . Accordingly, a simple equivalent resistance of R ds (Q 1 , Q 5 , Q 2 ) is shown by equivalent circuit  403  in  FIG. 4D . 
   Assume now that input terminals T 1  and T 2  are connected to a large positive overvoltage.  FIG. 5A  shows a schematic diagram for current limiter circuit  500  for large positive limiter voltage V L  conditions, such that V L &gt;&gt;+1 V. Because transistors Q 2  and Q 4  are of the depletion-mode MOSFET type, transistors Q 2  and Q 4  are in their full ON state, thereby having a low resistance between their drain and source terminals. Consequently, transistors Q 2  and Q 4  can be replaced by equivalent low-value R ds  resistors.  FIG. 5B  shows a schematic diagram for an equivalent circuit  501  having transistors Q 2  and Q 4  replaced by low-value R ds  resistors. Transistor Q 3  and resistor R 3  form a current source I 1  that outputs a current determined by the VgsOff voltage of Q 3  and R 3 .  FIG. 5C  shows a schematic diagram for an equivalent circuit  502  having transistor Q 3  and resistor R 3  replaced by source I 1 . Resistor R 1  is in series with current source I 1  and, therefore, can be eliminated from the equivalent circuit. R ds  of transistor Q 4  can be approximated by a wire because resistors R 2  and R 4  are each greater than 10 kΩ and R ds  of transistor Q 4  is &lt;100 Ω. A bias voltage for the gate of Q 5  is then developed across R 2 +R 4  in conjunction with the current source I 1 , as shown by equivalent circuit  503  in  FIG. 5D . Transistor Q 1  forms a current source that outputs a current that is determined by its VgsOff voltage and the R ds  resistance of transistor Q 5 . The R ds  resistance value of transistor Q 5  is then defined by its gate voltage which is approximately I 1 *(R 2 +R 4 ). This voltage is designed to be greater than the VgsOff voltage of Q 5 , and therefore turns off transistor Q 5  and along with it the current flow through transistor Q 1 , as shown by equivalent circuit  504  in  FIG. 5E . Resistors R 2  and R 4  are in series with current source I 1  and, therefore, can be eliminated. Thus, the only active current path between terminals T 1  and T 2  is the current source I 1  for large positive limiter voltage V L  conditions, as shown by equivalent circuit  505  in  FIG. 5F . 
   The opposite overload condition of a large negative voltage is shown in the equivalent models of  FIGS. 6A–6F . Operation proceeds as similarly described for the positive overload case, but with the actions of the symmetrical devices reversed. Specifically,  FIG. 6A  shows a schematic diagram for current limiter circuit  600  for large negative limiter voltage V L  conditions, such that V L &lt;&lt;−1 V. Because transistors Q 1  and Q 3  are of the depletion-mode MOSFET type, transistors Q 1  and Q 3  are in their full ON state, thereby having a low resistance between their drain and source terminals. Consequently, transistors Q 1  and Q 3  can be replaced by equivalent low-value R ds  resistors.  FIG. 6B  shows a schematic diagram for an equivalent circuit  601  having transistors Q 1  and Q 3  replaced by low-value R ds  resistors. Transistor Q 4  and resistor R 4  form a current source I 2  that outputs a current determined by the VgsOff voltage of Q 4  and R 4 .  FIG. 6C  shows a schematic diagram for an equivalent circuit  602  having transistor Q 4  and resistor R 4  replaced by source I 2 . Resistor R 2  is in series with current source I 2  and, therefore, can be eliminated from the equivalent circuit. R ds  of transistor Q 3  can be approximated by a wire because resistors R 1  and R 3  are each greater than 10 kΩ and R ds  of transistor Q 3  is &lt;100 Ω. A bias voltage for the gate of Q 5  is then developed across R 1 +R 3  in conjunction with the current source I 2 , as shown by equivalent circuit  603  in  FIG. 6D . Transistor Q 2  forms a current source that outputs a current that is determined by its VgsOff voltage and the R ds  resistance of transistor Q 5 . The R ds  resistance value of transistor Q 5  is then defined by its gate voltage which is approximately I 2 *(R 1 +R 3 ). This voltage is designed to be greater than the VgsOff voltage of transistor Q 5 , and therefore turns off transistor Q 5  and along with it the current flow through transistor Q 2 , as shown by equivalent circuit  604  in  FIG. 6E . Resistors R 1  and R 3  are in series with current source I 2  and, therefore, can be eliminated. Thus, the only active current path between terminals T 1  and T 2  is the current source I 2  for large negative limiter voltage V L  conditions, as shown by equivalent circuit  605  in  FIG. 6F . 
   Transistors Q 1 –Q 4  are high voltage N-channel depletion-mode MOSFETs. Transistors Q 1 –Q 4  provide blocking capability of many hundreds of volts, and can easily be cascaded for blocking thousands of volts. Transistor Q 5  is a low-voltage P-channel JFET that operates as a variable resistor. Because the VgsOff of transistor Q 5  may be greater than the VgsOff of transistors Q 3  and Q 4 , resistors R 1  and R 2  are needed for producing the required gate voltage for transistor Q 5 . The values of resistors R 1 –R 4  are selected for controlling the operating characteristics of current limiter circuit  300 . 
     FIG. 7  shows an exemplary graph illustrating the three operating regions of current limiter circuit  300 . The first operating region is a constant resistance region in which current limiter circuit  300  can be characterized by a constant resistance. When operating in the constant resistance region, current through current limiter circuit  300  increases proportionally with increasing voltage in the same manner as a constant resistance. The second operating region is a transition region in which the operating characteristics of current limiter circuit  300  transitions from a constant resistance region to a constant current region. The third operating region is a constant current region in which current through current limiter circuit  300  remains substantially constant for increasing voltage across the limiter. 
     FIG. 8  is a graph illustrating current as a function of voltage across current limiter circuit  300  with respect to typical input characteristics for other conventional current limiting devices. The current vs. voltage characteristics of current limiter circuit  300  are shown by curve  801 . At low voltage, current limiter circuit  300  exhibits a linear resistance of about 33 Ω having a thermal noise of about 0.75 nV/RtHz. For voltages greater than about 25 V, the current is limited to a constant 200 μA. A maximum current of about 50 mA flows at about 2 V. Between about 2 V and about 25 V, current limiter circuit  300  transitions between a constant resistance region and a constant current source region. 
   The current vs. voltage characteristics for the conventional input protection circuit of U.S. Pat. No. 5,742,463 to Harris are shown by curve  802 . The Harris input protection circuit exhibits a breakdown voltage of about 30 V because the entire differential limiter voltage appears on the gates of the transistors. In contrast, current limiter circuit  300  operates easily to the full source-drain breakdown voltage of the transistors, extending to many hundreds of volts. Moreover, the voltage blocking capability of the present invention can be increased into the thousands of volts by cascading transistors. 
   Other curves representing current vs. voltage characteristics that are shown in  FIG. 8  include curve  803  for a PTC thermistor having a resistance of 18 Ω and a thermal noise of 0.55 nV/RtHz; curve  804  for a light bulb having a resistance of 560 Ω and a thermal noise of 3.1 nV/RtHz; and curve  805  for a 1 kΩ resistor having a thermal noise of 4.1 nV/RtHz. 
     FIG. 9  is a graph illustrating power dissipation as a function of voltage across current limiter circuit  300  with respect to typical input characteristics for other conventional current limiting devices. Curve  901  shows that the power dissipation of current limiter circuit  300  is below 250 mW at any input voltage up to about 1000 V. In contrast, the power dissipation the conventional input protection circuit of U.S. Pat. No. 5,742,463 to Harris is shown by curve  902 .  FIG. 9  also shows other curves representing typical power dissipation as a function of limiter voltage. Curve  903  is the typical power dissipation for a PTC thermistor. Curve  904  is the typical power dissipation for a light bulb having a resistance of 560 Ω. Lastly, curve  905  is the typical power dissipation for a 1 kΩ resistor. 
     FIG. 10  is a graph illustrating nonlinearity characteristics as a function of voltage across current limiter circuit  300  with respect to the conventional input protection circuit of U.S. Pat. No. 5,742,463 to Harris. Curve  1001  shows the nonlinearity characteristics of current limiter circuit  300 , and curve  1002  shows the nonlinearity characteristics of the conventional input protection circuit of U.S. Pat. No. 5,742,463 to Harris. Current limiter circuit  300  provides a lower distortion in the normal operating range in comparison to the conventional Harris input protection circuit. At some voltages, the distortion exhibited by current limiter circuit  300  is better than the distortion exhibited by the Harris input protection circuit by an order of magnitude. 
     FIG. 11  is a graph illustrating inrush current characteristics as a function of a 100 Volt step across current limiter circuit  300  with respect to typical input characteristics for other conventional current limiting devices. Curve  1101  shows the inrush current characteristics for current limiter circuit  300 . Current limiter circuit  300  has about a 50 mA narrow transient (about 100 nS in duration) and then holds a constant current of about 200 μA thereafter. Accordingly, the requirements and stress placed on any clamp diodes coupled to output terminal T 2  are significantly reduced. Current limiter circuit  300  has about two orders of magnitude less inrush current than a PTC thermistor, as shown by curve  1103 . Curve  1104  shows the inrush current characteristics for a light bulb having a resistance of 560 Ω. Curve  1105  shows the inrush current characteristics for a 1 kΩ resistor. 
   Although the foregoing invention has been described in some detail for purposes of clarity of understanding, it will be apparent that certain changes and modifications may be practiced that are within the scope of the appended claims. Accordingly, the present embodiments are to be considered as illustrative and not restrictive, and the invention is not to be limited to the details given herein, but may be modified within the scope and equivalents of the appended claims.