Abstract:
A closed loop gain circuit controls the gain of a variable gain amplifier and provides a stable AGC response irrespective of the actual gain level. The amplifier may be arranged to amplify electrical signals output from a fiber optic gyroscope. A perturbation injection circuit provides a perturbation signal ±d to a phase modulator connected to the fiber optic gyro. A perturbation compensation circuit applies perturbation compensation signals to signals output from the variable gain amplifier and produces a compensated signal by reducing the magnitude of the perturbation in the amplified signal output from the variable gain amplifier. A gain error circuit connected to the perturbation compensation circuit produces a gain error signal that indicates the magnitude of the perturbation signal remaining in the amplified signal after perturbation compensation. A feedback system provides a gain control signal to the variable gain amplifier to reduce the magnitude of the gain error signal.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
     Applicants claim the benefit of U.S. Provisional Application Ser. No. 60/221,291, filed Jul. 27, 2000 for Broadband Acoustic Sensor. 
    
    
     BACKGROUND OF THE INVENTION 
     This invention relates generally to signal processing in fiber optic gyroscope systems. This invention relates particularly to automatic gain control (AGC) circuits in fiber optic gyroscope signal processing systems. Still more particularly, this invention relates to an AGC circuit that achieves a stable response irrespective of the actual gain level. 
     A closed-loop fiber optic gyroscope requires a high bandwidth, high performance signal processing scheme to capture the differential phase difference induced by the rotation rate. A description of the control loop is contained in U.S. Pat. No. 5,883,716, which issued to Mark and Tazartes on Mar. 16, 1999 and which is assigned to Litton Systems, Inc., assignee of the present invention. The disclosure of U.S. Pat. No. 5,883,716 is incorporated by reference into the present disclosure. In order to achieve the high degree of performance required while accommodating wide variations in loop gain, an active gain control scheme is required. Loop gain variations are due to aging of the light source, variations in optical output or loss over the operating temperature range, and temperature sensitivity of electro-optic components such as photodetectors. In addition, the optical signal can vary by a significant amount from instrument to instrument due to component and manufacturing tolerances. 
     Automatic gain control loops have therefore been utilized in the past to ensure that the total loop gain remains constant, which is essential to achieving maximum bandwidth and high order loop response. In the past, analog multipliers were used as gain stages in the detection path of the fiber optic gyroscope. While these devices provided an ideal linear control law (i.e. the gain is directly proportional to the applied control voltage), they exhibited a number of undesirable characteristics. These include bandwidth limitations, noise, cost, and linearity as a function of signal level. 
     For high performance, low noise fiber optic gyroscopes, an alternate gain control block was therefore considered. This is a variable gain amplifier whose gain in dB is proportional to the applied control voltage. In essence, this implies that the gain of the amplifier is an exponential function of applied control voltage as opposed to a linear function as in the earlier embodiments. Such devices are now readily available and offer lower noise and of course, a wider range of gain adjustment without substantial degradation in signal gain linearity. A gain range of 10:1 is easily achievable with such a device. 
     The desire to adapt such variable gain amplifiers in a fiber optic gyroscope circuit introduced a new problem which this invention addresses. Because of the non-linear gain control law, the time constant or response time of the AGC (automatic gain control) itself could be highly variable, thus limiting its ability to start-up rapidly and to track changes rapidly. 
     SUMMARY OF THE INVENTION 
     The present invention overcomes the deficiencies of the prior art by providing a stable AGC response irrespective of the actual gain level. 
     A closed loop gain control circuit according to the present invention for controlling the gain of a variable gain amplifier that is arranged to amplify electrical signals indicative of optical signal signals output from a fiber optic gyroscope comprises a perturbation injection circuit arranged to provide a perturbation signal ±d. A phase modulator is connected between the perturbation injection circuit and the fiber optic gyro. The phase modulator is arranged to apply the perturbation to the fiber optic gyroscope so that the perturbation signal is superimposed on the gyro output. A variable gain amplifier is arranged to receive the electrical signals indicative of optical signal signals output from the fiber optic gyroscope and provide an amplified signal. A perturbation compensation circuit is arranged to apply perturbation compensation signals to signals output from the variable gain amplifier. The perturbation compensation circuit produces a compensated signal by reducing the magnitude of the perturbation in the amplified signal output from the variable gain amplifier. A gain error circuit is connected to the perturbation compensation circuit. The gain error circuit produces a gain error signal that indicates the magnitude of the perturbation signal remaining in the amplified signal after perturbation compensation. A system processor is connected between the gain error circuit and the variable gain amplifier. The system processor provides a gain control signal to the variable gain amplifier to reduce the magnitude of the gain error signal. Processing circuitry is connected between the perturbation compensation circuit and the phase modulator for determining the rotation rate sensed by the fiber optic gyroscope and for controlling the phase modulator to apply a rate nulling signal to the fiber optic gyroscope. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a simplified block diagram of a fiber optic gyroscope system that includes automatic gain control circuitry according to the present invention; 
     FIG. 2 is a block diagram of the primary loop of the fiber optic gyroscope system of FIG. 1; and 
     FIG. 3 is a block diagram of an algorithm that may be included in the present invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring to FIG. 1, a fiber optic rotation sensor system  10  includes a fiber optic gyroscope  12  that includes a fiber optic sensing coil (not shown) that detects rotations about a sensing axis perpendicular to the plane of the sensing coil by means of the well-known Sagnac effect. The fiber optic gyroscope  12  produces an optical signal of fluctuating intensity determined by interference between counter-propagating waves in the sensing coil. The interference pattern indicates the rate of rotation of the sensing coil about its sensing axis. 
     The present invention includes a perturbation injection circuit  14  that provides a perturbation signal to a modulator  16 . The modulated perturbation signal is then input from the modulator  16  to the fiber optic gyroscope  12 . The modulated perturbation signal is superimposed on the optical signal produced by the fiber optic gyroscope  12  in accordance with the Sagnac effect. 
     The signal output from the fiber optic gyroscope  12  is incident upon a photodetector  18 , which produces an analog electrical signal that indicates the intensity of the optical output from the fiber optic gyroscope  12 . The electrical signal from the photodetector  18  is then amplified by a variable gain amplifier  20 . The amplified analog signal is input to an analog to digital (A/D) converter  22  that produces a digital signal that is used in further processing of the output of the fiber optic gyroscope  12 . 
     The digital signal output from the A/D converter  22  is input to a compensation circuit  24  that compensates for the injected perturbation. The output of the compensation circuit  24  is input to a rate demodulation processing circuit  26  and to a demodulator  28 . 
     The rate demodulation processing circuit  26  determines the rotation rate. A signal indicative of the rotation rate is output by the rate demodulation processing circuit  26  and input to a gyro control loop and modulation control circuit  30 . The fiber optic rotation sensor  10  is arranged in the well-known phase nulling configuration. Accordingly, the gyro control loop and modulation control circuit  30  provides a signal to the modulator  16  that nulls the phase shift in the sensing coil caused by the Sagnac effect. 
     The demodulator  28  detects errors in the gain of the variable gain amplifier  20 . A digital signal that indicates gain errors is input to a linearization and integration circuit  32 , which provides the linearized and integrated gain error signal to a digital to analog (D/A) converter  34 . The D/A converter  34  then provides the analog gain error signal to the gain control input of the variable gain amplifier  20 . 
     The AGC function is accomplished by injecting a know perturbation ±d in the loop and demodulating the resulting compensated signal with appropriate signs based on the polarity of the compensation. 
     The variable gain amplifier  20  used in the present invention employs a variable gain element whose gain in dB is proportional to the control voltage. Prior designs used a linear amplifier (not shown). The non-linear characteristic provided by the variable gain amplifier  20  is used advantageously in the gain control loop. 
     FIG. 2 is a more detailed block diagram of the AGC system according to the present invention. The fiber optic gyro  12  is responsive to the sum of the Sagnac phase shift and the phase shift produced by the modulator  16 . A summing junction  40  receives a signal input from the phase modulator  16  and a signal (SSF)Ω that represents the phase shift induced by the angular rate Ω where SSF is the Sagnac scale factor of the fiber gyro  12 . The summing junction  40  adds the signal received from the phase modulator  16  by the scale factor to produce a signal that is input to the fiber gyro  12 . The fiber gyro  12  uses the Sagnac effect to produce a signal            I   0     2        sin                   φ   M                            
     where φ M  is the phase difference between the modulated counter-propagating waves in the sensing coil. The optical signal output of the fiber gyro  12  is converted to an electrical signal by the photodetector  18 . The photodetector  18  has a scale factor K pd  which relates the electrical current output from the photodetector  18  to the optical power incident thereon. The photodetector typically has a scale factor K pd =0.9 A/W. 
     The photodetector output is then amplified by a transimpedance amplifier  42  that has a scale factor K TI . The transimpedance amplifier  42  is connected between the photodetector  18  and the variable gain amplifier  20  and serves to match the output impedance of the photodetector  18  to the input impedance of the variable gain amplifier  20 . The amplified electrical signal output from the variable gain amplifier  20  is input to a filter circuit  44 , which has scale factor k filt  that is preferably about 0.6. 
     The signal output from the filter circuit  44  is input to the A/D converter  22 , which converts the analog electrical signals into digital signals that are used for processing the fiber gyro output and for AGC of the variable gain amplifier  20 . The digital signal output from the D/A converter  22  is input to a scaling circuit  46 . The A/D converter  22  and the scaling circuit  46  together have a scale factor K A  that preferably is about 6554 bits/volt. 
     The output of the scaling circuit  46  is input to a summing junction  24  which also receives an input signal ±rd that is used for dither compensation. Signals output from the summing junction  24  are input to a rate loop control circuit  50  and to an AGC demodulator  52 . 
     The rate loop control circuit  50  may include one or more integrators that operate on the signals. U.S. Pat. No. 5,883,716, which issued to Mark and Tazartes on Mar. 16, 1999 and assigned to Litton Systems, Inc., assignee of the present invention discloses a rate loop control circuit that may used as the rate loop control circuit in the present invention. The output of the rate loop control circuit  50  is input to a digital gain circuit  64  that applies a gain of 2 s  to the output of the rate loop controller  50 . The output of the digital gain circuit  64  is input to a summing junction  66 , which applies the modulation by the dither signal ±d. After the dither is introduced into the circuit, the output of the summing junction  66  is input to a summing circuit  68  that produces a ramp signal output that is input to a scaling circuit  70 . The scaling circuit  70  multiplies the ramp signal by 2 −16  and provides an output signal to a multiplier  72  that also receives a signal to indicate the phase modulator scale factor PMSF. Signals output from the multiplier  72  are input to a D/A converter  76 , which converts the digital signal input thereto into an analog signal suitable for input to the phase modulator  16 . 
     The dither compensated signal input to the AGC demodulator  52  is demodulated with the appropriate signs based on the polarity of the compensation. The demodulated signal is then input to an integrator  78  which in turn provides an output to a system processor  80 . Ideally, if the forward gain of the fiber optic gyroscope signal path is correct, the compensation ±rd will exactly cancel the signal generated by the perturbation, and the demodulated value will be zero. If, however, the gain is in error, a residual signal will survive, which results in a non-zero demodulated value. The sign and magnitude of the demodulated value are indicative of whether the gain is high or low and by how much. The system processor  80  produces a gain control signal to adjust the gain accordingly to bring the residual signal to zero. The gain control signal from the system processor  80  is input to a gain control A/D converter,  82 , which, in turn, provides the analog gain control signal to the gain control input of the variable gain amplifier  20 . 
     The following mathematical analysis explains additional details of the method of operation of the AGC function provided by the present invention. The gain characteristic for the amplifier  20  is given by 
     
       
           G ( V )= g   0 10 αV ,  (1) 
       
     
     where α is a constant and V is the voltage applied to the gain control input of the amplifier  20 . 
     The gain expressed in dB is written as 
     
       
         20 log( G ( V ))=20 log( g   0 )+20 αV,   (2) 
       
     
     where g 0  is the normalizing gain, and 20α is the gain sensitivity in dB/volt. 
     Alternatively, the gain control law may be rewritten to relate to the binary control word driving the D/A converter  34 , which in turn generates the control voltage. Accordingly, the gain is given by 
     
       
           G ( b )= g   0 10 βb   (3) 
       
     
     where β is the digital control word written to the D/A converter  34         (     for                 example                   1   256          1   bit       )                          
     and b is the number of bits in the binary control word (for example between 0 and 255 bits). 
     The gain may also be expressed in dB using the following expression: 
     
       
         20 log( G ( b ))=20 log( g   0 )+20 βb,   (4) 
       
     
     where 20β is the gain sensitivity in dB/bit. It should be noted that β/α is the AGC DAC scale factor in volts/bit. 
     Following the loop in FIG. 2 from the point of injection of the perturbation ±d to the demodulator  28  where the AGC error is detected yields (after demodulation)              D   =     rd   -       π     2   16              I   0     2        sin                   φ   M          K   PD          K   TI          GK   FILT          K   A          d   .                 (   5   )                                
     The gain G 0 =G(b 0 ) that satisfies the overall loop gain is defined by the following relationships:                G   L     =       π     2   31              I   0     2        sin                   φ   M          K   PD          K   TI          G   0          K   FILT            K   A     ·     2   S                 (   6   )                                
     and 
     
       
           r=G   L ·2 −S .  (7) 
       
     
     where G L  is the overall desired loop gain. Thus the following expressions are obtained:                D   =       (         G   L     ·     2     -   S         -         G   L         G   0     ·     2   S              G        (   b   )           )        d            
          and             (   8   )               D   =       2     -   S              G   L          (     1   -       G        (   b   )         G   0         )            d   .               (   9   )                                
     Define the quantity G(b) as 
     
       
           G ( b )= G   0 (1−ε),  (10) 
       
     
     where ε is the gain error. Using Eq. (10) in Eq. (9) then gives the result that 
     
       
           D =2 −s   G   L   εd.   (11) 
       
     
     In actuality the AGC error signal is summed over many gyro transit times. The gyro transit time is the time interval required for an optical signal to propagate through the length of the sensing coil. The resulting summation is written as:                ∑   D     =         Δ                 T     τ          2     -   S            G   L        ɛ                 d             (   12   )                                
     where ΔT is the integration time and τ is the transit time. The estimate of the relative gain error is then given by                ɛ   ^     =       τ     Δ                 T              2   S         G   L        d            ∑     D   .                 (   13   )                                
     The gain control law is expressed as                1   -   ɛ     =         G        (   b   )         G   0       =         G        (   b   )         G        (     b   0     )         =       10     β        (     b   -     b   0       )         =     10     β                 Δ                 b                     (   14   )                                
     where Δb is the error in the gain control D/A converter  34 . 
     Taking the natural logarithm of Eq. (14) gives 
     
       
         1 n (1−ε)=βΔ b 1 n (10).  (15) 
       
     
     Solving Eq. (15) for Δb gives                Δ                 b     =         ln        (     1   -   ɛ     )         β                   ln        (   10   )           .             (   16   )                                
     Eq. (16) may be differentiated with respect to ε to yield:                    ∂   Δ                   b       ∂   ɛ       =       -     1     β                   ln        (   10   )                    1     1   -   ɛ       .               (   17   )                                
     Eq. (17) is used to form the linearized update equation:                    Δ                 b     ≈           ∂   Δ                   b       ∂   ɛ            ɛ   ^         =       -     1     β                   ln        (   10   )                    ɛ   ^       1   -   ɛ           ,           (   18   )                                
     which is rearranged to yield                  b        (     n   -   1     )       =         b        (   n   )       -         Δ                 T       t   c          Δ                 b       =       b        (   n   )       +         Δ                 T       t   c            1     β                   ln        (   10   )                  ɛ   ^       1   -     ɛ   ^                 ,           (   19   )                                
     where t c  is the desired time constant for the AGC loop. In order to ensure stability of the above equations, the value of {circumflex over (ε)} should be limited in accordance with the gain range. For a 10 to 1 range, the limits are −9.0≦{circumflex over (ε)}≦0.9. 
     FIG. 3 illustrates an algorithm of signal processing that may be used to control the gain of the amplifier  20 . At initialization:                κ   ɛ     =       τ     Δ                 T              2   S         G   L        d                 (   20   )                                
     where 
     τ is the loop transit time; 
     ΔT is the demodulator integration time; 
     s is the loop controller shift count; 
     G L  is the primary loop gain (1.0 for deadbeat control, 0.2 for integral control); and 
     d is the dither perturbation value.                κ   AGC     =         Δ                 T       t   c            1     β                   ln        (   10   )                     (   21   )                                
     where t c  is the desired AGC loop time constant and (20)β is the variable gain amplifier sensitivity in dB/bit. 
     The gain control command b is set to a value 
       b=b   init   (22) 
     where b init  is the initial gain control D/A converter  34  command to be read from memory. At the integration interval (i.e., every ΔT), 
     
       
         ε= K   ε   ΣD   (23) 
       
     
     where ΣD is the demodulator output integrated over the interval ΔT. The limit of the estimated gain error ε is limited to [−9.0, 0.9]. The digital gain control command b supplied to the gain control D/A converter  34  may then be written as              b   =     b   -       κ   AGC                       ɛ     1   -   ɛ       .                 (   24   )                                
     Nominal values for parameters used in the algorithm are given in the following table. 
     
       
         
               
               
             
           
               
                   
               
             
             
               
                 τ 
                 5.4 μs 
               
               
                 ΔT 
                 500 μsec-1 sec 
               
               
                 s 
                 5 to 16 
               
               
                 G L   
                 0.1-1.0 
               
               
                 d 
                 2 16  to 2 28   
               
               
                 t c   
                 1.0 msec-30 sec 
               
               
                   
               
               
                 20β 
                 
                   
                     
                       
                         
                           20 
                            
                           
                               
                           
                            
                           β 
                         
                         = 
                         
                           
                             20 
                             256 
                           
                            
                           
                               
                           
                            
                           
                             dB 
                             / 
                             bit 
                           
                         
                       
                     
                             
                     
                         
                     
                   
                 
               
               
                   
               
               
                 β 
                 
                   
                     
                       
                         β 
                         = 
                         
                           1 
                           256