Abstract:
Clock recovery method for bursty communications. A method is disclosed for recovering the clock from a received data stream that comprising bursts of data with zones of substantially no data between the bursts of data. A receive clock is provided that operates within a reference frequency range. The time between data transitions in the received data is then measuring relative to the receive clock. A determination is then made if the measured time is substantially an integral of the receive clock. If not a substantial integral of the receive clock, the frequency of the receive clock is adjusted to compensate for the difference.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS  
       [0001]     This is a Continuation Application of U. S. Ser. No. 10/244,728, filed Sep. 16, 2002, entitled “CLOCK RECOVERY METHOD FOR BURSTY COMMUNICATIONS” (Atty. Dkt. No. CYGL-26,068) and it is related to U.S. patent application Ser. No. 09/885,459, filed Jun. 19, 2001 and entitled “FIELD PROGRAMMABLE MIXED-SIGNAL INTEGRATED CIRCUIT” (Atty. Dkt. No. CYGL-25,768 ), which is incorporated herein by reference and is co-pending of even date hereof with U.S. patent application Ser. No. 09/885,459, entitled “PRECISION OSCILLATOR FOR AN ASYNCHRONOUS TRANSMISSION SYSTEM,” and Ser. No. 10/244,344 (Atty. Dkt. CYGL-26,116), which is also incorporated herein by reference. 
     
    
     TECHNICAL FIELD OF THE INVENTION  
       [0002]     This invention pertains in general to clock recovery methods and, more particularly, a method for recovering the clock in a communications system having Bursty signal transitions, e.g., USB.  
       BACKGROUND OF THE INVENTION  
       [0003]     Serial bus communication protocols have long been utilized for communications between two devices. This serial communication can provide long range or short range communication between the two devices and can either be “synchronous” or “asynchronous.” For asynchronous transmission, there are provided two independent clocks, one at the master and one at the slave node (note that either device on either end of the communication path can be either the master or the slave) that are each operable to receive or transmit data based solely upon their clock. Asynchronous communication tends to be somewhat slower than synchronous communication since there will naturally be a finite error between the two clocks. For synchronous communication, either a separate clock signal is provided between the two devices on a separate clock line, or some type of clock recovery is utilized. One type of synchronous serial transmission protocol that utilizes a separate clock line is referred to as I 2 C. In a clock recovery system, the clock signal is overlapped with the data on the same line, such that the clock information can be recovered from data transitions. One type of such clock recovery protocol is Manchester coded PSK. Another type, that associated with the present disclosure, is Universal Serial Bus (USB).  
         [0004]     In order to maintain sync between the two systems, the receiver will typically “lock” onto the received data and extract the clock information therefrom. There will typically be provided a receive clock, which will have the frequency and phase thereof varied to substantially equal the frequency and phase of a transmit clock which is extracted from the receive data. One technique for providing this receive clock and adjusting the frequency and phase thereof is a phase locked loop. For continuous transmission systems, such as Manchester coded PSK, data transmissions are present on a substantially continual base, such that the phase and frequency error between the receive clock and the transmit clock and be continually minimized or corrected for. However, with respect to the USB transmission system, these have what is referred to as “bursty” communications; that is, data is only present in bursts. Therefore, substantially continual data transitions are not present in order for a phase locked loop to lock onto. As such, during times of no transmission, the receive clock may drift in phase and frequency and, upon receipt of the next burst of data, lock will again have to be acquired before the integrity of the data reception can be guaranteed.  
       SUMMARY OF THE INVENTION  
       [0005]     The present invention disclosed and claimed herein, in one aspect thereof, comprises a method for recovering the clock from a received data stream comprising bursts of data with zones of substantially no data between the bursts of data. A receive clock is provided that operates within a reference frequency range. The time between data transitions in the received data is then measuring relative to the receive clock. A determination is then made if the measured time is substantially an integral of the receive clock. If not a substantial integral of the receive clock, the frequency of the receive clock is adjusted to compensate for the difference.  
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0006]     For a more complete understanding of the present invention and the advantages thereof, reference is now made to the following description taken in conjunction with the accompanying Drawings in which:  
         [0007]      FIG. 1  illustrates an overall block diagram of a mixed-signal integrated circuit utilizing a USB port;  
         [0008]      FIG. 2  illustrates a more detailed diagram of the integrated circuit of  FIG. 1 ;  
         [0009]      FIG. 3  illustrates a block diagram of the UART;  
         [0010]      FIG. 3A  illustrates a block diagram of the baud rate generator;  
         [0011]      FIG. 4  illustrates a block diagram of the precision oscillator;  
         [0012]      FIG. 5  illustrates a more detailed diagram of the precision oscillator of  FIG. 4 ;  
         [0013]      FIG. 6  illustrates an output waveform diagram of a precision oscillator;  
         [0014]      FIG. 7  illustrates a schematic diagram of the temperature compensated reference voltage;  
         [0015]      FIG. 8  illustrates a schematic diagram of one-half of the output wave shaping circuit;  
         [0016]      FIG. 9  illustrates a schematic diagram/layout for one of the resistors illustrating the mask programmable feature thereof;  
         [0017]      FIG. 10  illustrates a schematic diagram of the programmable capacitor;  
         [0018]      FIG. 11  illustrates a schematic diagram of the comparator;  
         [0019]      FIG. 12  illustrates a logic diagram for the S/R latch in combination with the comparator;  
         [0020]      FIG. 13  illustrates a schematic diagram of the delay block;  
         [0021]      FIG. 14  illustrates a schematic diagram for an offset circuit for the comparator;  
         [0022]      FIG. 15  illustrates a block diagram of two computer peripheral devices;  
         [0023]      FIG. 16  illustrates the data stream for the USB port;  
         [0024]      FIG. 17  illustrates an overall block diagram for the system for recovering the clock from the bursty communication;  
         [0025]      FIG. 18  illustrates a diagrammatic view of the SP counter operation relative to the PH modulo-K count;  
         [0026]      FIG. 19  illustrates a diagram of the SP count value;  
         [0027]      FIG. 20  illustrates a diagrammatic view of the receive data transitions relative to the SP count;  
         [0028]      FIG. 21  illustrates a diagrammatic view of the operation of the SP counter and the BP counter;  
         [0029]      FIG. 22  illustrates an overall block diagram of the bursty communications clock recovery method;  
         [0030]      FIG. 23  illustrates a simplified flowchart depicting the basic control steps of the bursty communications method;  
         [0031]      FIG. 24  illustrates a block diagram of one instantiation of the oscillator; and  
         [0032]      FIGS. 25 and 26  illustrate tables for the oscillator controls.  
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0033]     Referring now to  FIG. 1 , there is illustrated an integrated circuit that is comprised of a fully integrated mixed-signal System on a Chip with a true 12-bit multi-channel ADC  110  with a programmable gain pre-amplifier s  12 , two 12-bit DACs  114  and  116 , two voltage comparators  118  and  120 , a voltage reference  22 , and an 8051 -compatible microcontroller core  124  with 32 kbytes of FLASH memory  126 . There is also provided an 12C/SMBUS  128 , a UART  130 , and an SPI  132  serial interface  140  implemented in hardware (not “bit-banged” in user software) as well as a Programmable Counter/Timer Array (PCA)  134  with five capture/compare modules. There are also 32 general purpose digital Port I/Os. The analog side further includes a multiplexer  113  as operable to interface eight analog inputs to the programmable amplifier  112  and to the ADC  110 .  
         [0034]     With an on-board V DD  monitor  136 , WDT, and clock oscillator  137 , the integrated circuit is a stand-alone System on a Chip. The MCU effectively configures and manages the analog and digital peripherals. The FLASH memory  126  can be reprogrammed even in-circuit, providing non-volatile data storage, and also allowing field upgrades of the 8051 firmware. The MCU can also individually shut down any or all of the peripherals to conserve power.  
         [0035]     A JTAG interface  142  allows the user to interface with the integrated circuit through a conventional set of JTAG inputs  144 . On-board JTAG debug support allows non-intrusive (uses no on-chip resources), full speed, in-circuit debug using the production integrated circuit installed in the final application. This debug system supports inspection and modification of memory and registers, setting breakpoints, watchpoints, single stepping, run and halt commands. All analog and digital peripherals are fully functional when debugging using JTAG.  
         [0036]     The microcontroller  140  is fully compatible with the MCS-51™ instruction set. Standard 803x/805x assemblers and compilers can be used to develop software. The core has all the peripherals included with a standard 8052, including three 16-bit counter/timers, a full-duplex UART, 256 bytes of internal RAM, 128 byte Special Function Register (SFR) address space, and four byte-wide I/O Ports. A Universal Serial Bus (USB) interface is provided with a controller  160  that interfaces with memory  162  (of which all or a portion may be on the integrated circuit with the controller  160 ) and a USB transceiver  164 . The transceiver  164  will interface with dedicated pins  166  to receive/transmit serial data. This data is referred to as “bursty communications.” 
         [0037]     Referring further to  FIG. 1 , the core  141  is interfaced through an internal BUS  150  to the various input/output blocks. A cross-bar switch  152  provides an interface between the UART  130 , SPI BUS  132 , etc., and the digital I/O output. This is a configurable interface.  
         [0038]     The core  140  employs a pipelined architecture that greatly increases its instruction throughput over the standard 8051 architecture. In a standard 8051, all instructions except for MUL and DIV take 12 or 24 system clock cycles to execute with a maximum system clock of 12 MHz. By contrast, the core  140  executes seventy percent (70%) of its instructions in one or two system clock cycles, with only four instructions taking more than four system clock cycles. The core  140  has a total of 109 instructions. The number of instructions versus the system clock cycles to execute them is as follows:  
                                                                                                                                   Instructions                26   50   5   14   7   3   1   2   1                        Clocks to Execute   1   2   ⅔   3   ¾   4   ⅘   5   8                  
 
         [0039]     With the core  140 &#39;s maximum system clock at 20 MHz, it has a peak throughput of 20 MIPS.  
         [0040]     As an overview to the system of  FIG. 1 , the cross-bar switch  152  can be configured to interface any of the ports of the I/O side thereof to any of the functional blocks  128 ,  130 ,  132 ,  134  or  136  which provide interface between the cross-bar switch  152  and the core  140 . Further, the cross-bar switch can also interface through these functional blocks  128 - 136  directly to the BUS  150 .  
         [0041]     Referring now to  FIG. 2 , there is illustrated a more detailed block diagram of the integrated circuit  FIG. 1 . In this embodiment, it can be seen that the cross-bar switch  152  actually interfaces to a system BUS  202  through the BUS  150 . The BUS  150  is a BUS as operable to allow core  140  to interface with the various functional blocks  128 - 134  in addition to a plurality of timers  204 ,  206 ,  208  and  210 , in addition to three latches  212 ,  214  and  216 . The cross-bar switch  152  is configured with a configuration block  220  that is configured by the core  140 . The other side of the cross-bar switch  152 , the I/O side, is interfaced with various port drivers  222 , which is controlled by a port latch  224  that interfaces with the BUS  150 . In addition, the core  140  is operable to configure the analog side with an analog interface configuration in control block  226 .  
         [0042]     The core  140  is controlled by a clock on a line  232 . The clock is selected from, as illustrated, one of two locations with a multiplexer  234 . The first is external oscillator circuit  137  and the second is an internal oscillator  236 . The internal oscillator circuit  236  is a precision temperature compensated oscillator, as will be described hereinbelow. The core  140  is also controlled by a reset input on a reset line  154 . The reset signal is also generated by the watchdog timer (WDT) circuit  136 , the clock and reset circuitry all controlled by clock and reset configuration block  240 , which is controlled by the core  140 . Therefore, it can be seen that the user can configure the system to operate with an external crystal oscillator or an internal precision non-crystal non-stabilized oscillator that is basically “free-running.” This oscillator  236 , as will be described hereinbelow, generates the timing for both the core  140  and for the UART  130  timing and is stable over temperature.  
         [0043]     Referring now to  FIG. 3 , there is illustrated a block diagram of the UART  130 . A system clock is input to a baud rated generator  302  which provides a transmit clock on the line  304  and a receive clock on a line  306 . The transmit clock is input to a transmit control block  308  and the receive clock is input to a receive control block  310 . A serial control register (SCON 0 )  320  is provided that is operable to provide control signals to the control blocks  308  and  310 . The transmit data is received from a bus  322  and is input through a gate  324  to a serial data buffer (SBUF)  326 . The output of this data is input to a zero detector  328  and then to a control block  308 . The system is an asynchronous, full duplex serial port device and two associated special function registers, a serial control register (SCON 0 )  320  and a serial data buffer (SBUF 0 ) (not shown), are provided. Data is received on a line  312  and is input to an input shift register  314 . This is controlled by the control block  310  to output the shifted-in data to a latch  332  and then through a gate  334  to an SFR bus  322 . In transmit mode, data is received from an SFR bus  321  and input through a gate  324  to a transmit shift register  326  which is output to a transmit line  319  from the register  326  or from the control block  308  through an AND gate  338  which is input to one input of an OR gate  340  to the transmit line  319 . This is all controlled by the control block  308 .  
         [0044]     Referring now to  FIG. 3A , there is illustrated a block diagram of the baud rate generator  302 . This baud rate is generated by a timer wherein a transmit clock is generated by a block TL 1  and the receive clock is generated by a copy of the TL 1  illustrated as an RX Timer, which copy of TL 1  is not user-accessible. Both the transmit and receive timer overflows are divided by two for the transmit clock and the receive clock baud rates. The receive timer runs when timer  1  is enabled, and uses the same TH 1  value, this being a reload value. However, an RX Timer reload is forced when Start Condition is detected on the receive pin. This allows a receipt to begin any time a Start is detected, independent of the state of the transmit timer.  
         [0045]     Referring now to  FIG. 4 , there is illustrated a diagrammatic view of the precision internal oscillator  236  that is disposed on integrated circuit. The integrated circuit, as noted hereinabove, is a commercially available integrated circuit that incorporates the precision oscillator  236  in association therewith. The integrated circuit provides the capability of selecting a crystal oscillator wherein a crystal is disposed between two crystal ports, selecting an external clock signal or selecting an internal free-running oscillator. The free-running oscillator is illustrated in  FIG. 4  as the precision oscillator  236 . At the center of the oscillator are two comparators, a first comparator  402  and a second comparator  404 . A temperature compensated voltage reference circuit  406  is provided that provides a temperature compensated voltage reference (the trip voltage V TRIP ) to the negative inputs of the comparators  402 . The outputs of the comparators  402  and  404  are connected to the Set and Reset, respectively, inputs of an S/R latch  408 . The Q and Q-Bar outputs thereof are input to an output RC timing circuit  410  that is operable to define the period of the oscillator, the output of the S/R latch  408  providing the output clock signal. The output of this RC timing circuit  410  is fed back to the positive inputs of the comparators  402  and  404 . The output RC timing circuit  410  is also temperature compensated. As will be described hereinbelow, the voltage reference block  406  provides a negative temperature coefficient, whereas the comparators  402  and S/R latch  408  combination provide a positive temperature coefficient and the output RC timing circuit  410  provide a positive temperature coefficient. The overall combined coefficient will be approximately zero, as will be described hereinbelow.  
         [0046]     Referring now to  FIG. 5 , there is illustrated a more detailed diagrammatic view of the precision oscillator of  FIG. 4 . The voltage reference circuit  406  is comprised of a voltage divider that divides the supply voltage V DD  to a voltage V TRIP  on a node  502 . The voltage divider is comprised of a top resistor  504  labeled R 3 . The bottom half of the voltage divider is comprised of two parallel resistors, a resistor  506  labeled R 2  and a resistor  508  labeled R 4 . For nomenclature purposes, the resistors will be referred as R 2 , R 3  and R 4 .  
         [0047]     Resistors R 3  and R 4  are fabricated from the same material to provide a positive temperature coefficient. These are fabricated from the N-diffusion material, which has a positive temperature coefficient. By comparison, R 2  is manufactured from polycrystalline silicon in the first layer which is referred to as Poly 1  material, and which also has a positive temperature coefficient, but which differs. It should be understood that different materials could be utilized, it only being necessary that there be two resistors having different temperature coefficients. Although not a part of this disclosure, Poly 1  material is basically the first layer of polycrystalline silicon that is disposed on the substrate over a protective oxide layer, from which such structures as the gates of transistors are fabricated. With the positive temperature coefficients of the resistors, this will result in the voltage V TRIP  having a negative coefficient. As will be described hereinbelow, the resistors being of different materials facilitates adjustments between the two resistors R 2  and R 4  to vary the temperature coefficient. This is primarily due to the fact that they are of differing materials.  
         [0048]     The output RC timing circuit  410  is comprised of two RC circuits. The first RC circuit is comprised of a P-channel transistor  520  having the source/drain path thereof connected between V DD  and one side of a resistor  522  labeled R, the other end thereof connected to a node  524 . Node  524  is connected to one side of a capacitor  526 , the other side of the capacitor  526  connected to V SS . -channel transistor  528  has the source/drain path thereof connected across capacitor  526 , and the gate thereof connected to the gate of P-channel transistor  520  and also to the Q-output of the S/R latch  408 . Node  524  comprises the positive input of the comparator  402 . The second RC network is comprised of a P-channel transistor  530  having the source/drain path thereof connected between V DD  and one side of a resistor  532  (labeled R), the other side of resistor  532  connected to a node  534 . Node  534  is connected to one side of a capacitor  536 , the other side thereof connected to V SS . An N-channel transistor  538  has the source/drain path thereof connected between node  534  and V SS . The gate of transistor  538  is connected to the gate of transistor  530  and also to the Q-Bar output of S/R latch  408 . The node  534  comprises the positive input of the comparator  404 . The output waveform for the circuit of  FIG. 5  is illustrated in  FIG. 6 , wherein conventional RC rise and fall curves are illustrated for each of the RC circuits. The period of each output waveform is defined from the initial turn-on point where voltage is applied to the resistor R to the point where resistor R of the other of the RC circuits is turned on. There will be period T 1  and a period T 2  for each of the RC circuits, respectively. The sum of the two periods is equal to the period for the oscillator. Transistors  520 ,  530 ,  528  and  538  are sized such that their resistances are substantially less than the value of resistors  522  and  532 . The resistors  522  and  532  are fabricated from Poly 1  material due to its low temperature coefficient. The period of the oscillator is the sum of the period T 1  and the period T 2  plus two times the delay of the comparators.  
         [0049]     Referring now to  FIG. 7 , there is illustrated more detailed block diagram of the implementation of the voltage reference  406 . The resistor  504  which is illustrated in  FIG. 5  as being connected to V DD  is actually connected through the source/drain of the P-channel resistor  702  to V DD  with the gate thereof connected to a bias voltage. Similarly, the bottom end of resistor  506  is connected to V SS  through the source/drain path of a N-channel transistor  706  to V SS , the gates of both transistors  704  and  706  connected to a bias. Transistors  702 ,  704  and  706  are sized such that their resistances are substantially less than the value of resistors R 2 , R 3  and R 4 . Also, first order power supply independence comes from the fact that the trip voltage V Trip  is proportional to the supply voltage, i.e., V DD *(1−e(t/τ)). Therefore, in the time it takes to reach the trip voltage at the input of the comparator is supply independent to the first order. This is one reason that the RC timing circuits are utilized rather than a current source charging a capacitor, which does not provide the first order cancellation. 
 
 V   Trip   =V   DD * ratio 
 
 V   Trip   =V   DD *(1− e (− T 1/τ)) 
 
 T 1=−τ*1 n (1 −V   Trip   /V   DD ) 
 
Thus:  T 1=−τ*1 n (1−ratio) 
 
         [0050]     From a temperature compensation standpoint, there are a number of aspects of the voltage reference circuit  406  that can be utilized to provide temperature compensation. Commonly, the resistors have a set variation with respect to temperature. The Poly 1  resistor R 2  has a temperature coefficient of 255 ppm whereas the N-diffused resistors R 3  and R 4  have a temperature coefficient of 800 ppm. In the present disclosure, it is desirable to have a negative coefficient of 462 ppm.  
         [0051]     To analyze how a negative temperature coefficient is created with the resistors R 2 , R 3  and R 4 , consider that R 2  and R 4  are a parallel combination defined as REQ=R 2 //R 4 . If REQ and R 3  have different temperature coefficients with TCR 3 &gt;TCREQ, then the trip voltage will have a negative temperature coefficient. V TRIP  will be defined as follows:  
         V   TRIP     =       REQ       R   3     +   REQ       ⁢     V   DD           
           1     V   TRIP       ⁢       ⅆ     V   TRIP         ⅆ   T         =         1   REQ     ⁢       ⅆ   REQ       ⅆ   T         -         R   3         R   3     +   REQ       ⁡     [       1   REQ     ⁢       ⅆ   REQ       ⅆ   T         ]       -         R   3         R   3     +   REQ       ⁡     [       1     R   3       ⁢       ⅆ     R   3         ⅆ   T         ]             
           1     V   TRIP       ⁢       ⅆ     V   TRIP         ⅆ   T         =         R   2         R   3     +   REQ       ⁡     [     TCREQ   -     TCR   3       ]           
         V   TRIP     =       REQ       R   3     +   REQ       ⁢     V   DD           
 
         [0052]     For REQ, is must be assumed that V TRIP  is a fixed value, such that R 2  and R 4  can be varied to target a specific temperature coefficient. This can be shown by the following equations:  
           1   REQ     ⁢       ⅆ   REQ       ⅆ   T         =       [       1     R   2       ⁢       ⅆ     R   2         ⅆ   T         ]     +     [       1     R   4       ⁢       ⅆ     R   4         ⅆ   T         ]     -         R   2         R   2     +     R   4         ⁡     [       1     R   2       ⁢       ⅆ     R   2         ⅆ   T         ]       -         R   4         R   2     +     R   4         ⁡     [       1     R   4       ⁢       ⅆ     R   4         ⅆ   T         ]             
       TCREQ   =       TCR   2     +     TCR   4     -         R   2         R   2     +     R   4         ⁢     TCR   2       -         R   4         R   2     +     R   4         ⁢     TCR   4             
 
         [0053]     The results of equation 5 can be utilized in equation 3 to set the final temperature coefficient of V TRIP .  
         [0054]     Referring now to  FIG. 8 , there is illustrated a detailed diagram of the implementation of one-half of the charging structure  410 . This, as with the case with respect to the voltage reference structure  406 , there is provided a P-channel transistor  802  for connecting the top end of the resistor  522  to V DD , with the gate thereof connected to a bias supply. This P-channel transistor introduces very little error in the temperature operation thereof. Capacitor  526  is a variable capacitor, such that the value thereof can be varied to set the period for the oscillator. The capacitor  526  is fabricated from an insulator disposed between the first layer poly, P 1 , and the second layer poly, P 2 , with a layer of oxide disposed therebetween. The resistor  522  is an N-diffusion resistor.  
         [0055]     The resistors R 3 , R 2  and R 4  in the voltage reference circuit  406  are variable resistors that can be mask programmable resistors. Resistor R 3  is utilized to set the value of V TRIP  and resistors R 2  and R 4  are utilized to select a temperature coefficient, since they have dissimilar temperature coefficients.  
         [0056]      FIG. 9  illustrates a layout for one of the resistors R 2 -R 4 . A plurality of series connected resistors are provided that are fabricated in either the substrate with an N-type diffusion or in the Poly 1  layer. These resistors provide a mask programmable set of connections  904  to allow one or more resistors  902  to be added into the resistor string, they being initially shorted out. Although not shown, there is also provided the ability to short additional ones of the resistors to decrease the value. This is mask programmable and is utilized to “tweak” the design at the metal level.  
         [0057]     Referring now to  FIG. 10 , there is illustrated a diagrammatic view of the capacitor  526 , which is a register programmable capacitor to allow for adjustment of the center frequency. There is provided a nominal capacitor  1002  which has a value of 380 fF, which is connected between node  24  and V SS . In parallel therewith, there is also provided a mask programmable capacitor  1004  that provides for eight steps of programming in increments of 39.5 fF. The register programmable capacitors are provided with a capacitor  1006  of value “C” that is connected between a node  524  and one side of the source/drain path of an N-channel transistor  1008 , the gate thereof connected to the LSB bit. The configuration of the capacitor  1006  disposed between the switching transistor  1008  and the node  524  is only used for LSB. This structure allows the use of the smaller unit capacitor, but there is some non-linear capacitance that is introduced from the source/drain of the transistor  1008  and, also, the wire bonds. The remaining selectable capacitors are each comprised of a capacitor  1010  which is connected between V SS  and one side of the source/drain path of an N-channel transistor  1012 , the other side thereof connected to node  524  and the gate thereof connected to the bits [ 1 ] through [ 6 ]. The value of the capacitor  1010  associated with bit &lt; 1 &gt;is a value of “C”, with the next selectable capacitor  1010  having the associated transistor gate connected to the bit value &lt; 2 &gt;and the last of the selectable capacitor  1010  having the gate of the associated transistor connected to the bit &lt; 6 &gt;and a value of 32 C. This is a binary tree, with the LSB providing an LSB of approximately C/2.  
         [0058]     Referring now to  FIG. 11 , there is illustrated a diagrammatic view of the differential input structure for each of the comparators  402  and  404 . There are provided two differential P-channel transistors  1102  and  1104  having one side of the source/drain paths thereof connected to a node  1106 , node  1106  connected through a current source  1108  to V DD . The other side of the source/drain path of transistor  1102  is connected to a node  1110  and the other side of the source/drain path of transistor  1104  is connected to a node  1112 . The gate of transistor  1102  comprises the positive input and the gate of transistor  1104  comprises the negative input connected to V REF . Node  1110  is connected to one side of the source/drain path of an N-channel transistor  1114  and the gate thereof, the other side of the source/drain path of transistor  1114  connected to V SS . Node  1112  is connected to one side of the source/drain path of an N-channel transistor  1116 , the other side thereof connected to V SS  and the gate thereof connected to a node  1118 , node  1118  connected to one side of a resistor  1120 , the other side thereof connected to the gate of transistor  1114 . Node  1112  is also connected to the gate of an N-channel transistor  1122 , the source/drain path thereof connected between node  1118  and V SS . This structure is referred to as a modified Flynn-Lidholm latching comparator which provides a Set/Reset latch with dynamic logic, described in Flynn M. Lidholm S. U., “A 1.2 μm CMOS Current Controlled Oscillator, IEEE Journal of Solid state Circuits,” Vol. 27 No. 7 July 1992.  
         [0059]     Referring now to  FIG. 12 , there is illustrated a diagrammatic view of the comparator  402  and one-half of the S/R latch  408  illustrating the Q-Bar output. The one-half of the S/R latch  408  has the Set input thereof connected to the output of comparator  402  and input to the gate of an N-channel transistor  1202 , the source/drain path thereof connected between a node  1204  and V SS . A P-channel transistor  1206  has the source/drain path thereof connected between node  1204  and V DD , the gate thereof connected to a node  1208 . Node  1204  is connected to the input of a conventional inverter  1210  and also to one side of the source/drain path of an N-channel transistor  1212 , the other side thereof connected to V DD  and the gate thereof connected to a node  1214 , which node  1214  is also connected to the output of inverter  1210 . Node  1214  is connected to the input of an inverter  1216 , the output thereof providing the Q-Bar output. Node  1214  also is connected through a delay block  1218  to the input of a NAND gate  1220  labeled “ND1.” NAND gate  1220  is comprised of a P-channel transistor  1222  having the source/drain path thereof connected between V SS  and the node  1208  and an N-channel transistor  1224  having the source/drain path thereof connected between the node  1204  and one side of the source/drain path of an N-channel transistor  1226 , the other side thereof connected to V SS . The gates of transistors  1222  and  1224  are connected to the output of the delay block  1218 . The gate of transistor  1226  is connected to the reset input “RST” from the other side of the S/R latch  408 . Node  1208  is connected to the input of an inverter  1230 , the output thereof driving the gate of an N-channel transistor  1232  having the source/drain path thereof connected between the output of the comparator  402 , the SET input of latch  408 , and the other side of the source/drain path of transistor  1232  connected to V SS . The parallel structure to that associated with the output of comparator  402  in  FIG. 12  is provided for the output of comparator  404  for the Reset input.  
         [0060]     In operation, when the positive input of comparator  402 , FB 1 , charges up, SET starts to go high. As it reaches the threshold voltage V TH  of transistor  1202 , Q-Bar begins to go low and, at the same time, the other side of the latch, which has a NAND gate ND 2  similar to ND  1 , begins to go low and pulls down RST. When RST is pulled down, this then sets the Q-output. Initially, it is assumed that Q-Bar is set to a value of “1 ” and the Q-output is set to “0”with FB 1  equaling “0” on comparator  402  and FB 2  on the positive input of comparator  404  being initially set to “1 ” with SET=0 and RST=1. The delay block  1218  prevents ND 1  from pulling down the SET value before RST goes low. RST going low ensures that the pull down input is low (or ND 1  high) to result in a symmetric process for SET/RST.  
         [0061]     Referring now to  FIG. 13 , there is illustrated a schematic diagram of the delay block  1218 . This delay block is comprised of a plurality of series connected inverters comprised of two series connected transistors, a P-channel transistor  1302  and an N-channel transistor  1304 , with the gates thereof connected together and one side of the source/drain path thereof connected to a node  1306 , transistor  1302  connected between V DD  and V SS .  
         [0062]     Referring now to  FIG. 14 , there is illustrated a diagrammatic view of a simplified comparator illustrating how supply independence is enhanced. The comparator of  FIG. 14  is illustrated with a current source  1402  disposed between V DD  and a node  1404 , node  1404  connected to one side of two differential connected P-channel transistors  1406  and  1408 . The gate of transistor  1406  is connected to one input, whereas the gate of transistor  1408  is connected to the other V REF  input. The other side of the source/drain path of transistor  1406  is connected to a node  1410 , which is connected to one side of the source/drain path of an N-channel  1412 , the other side thereof connected to ground and the gate thereof connected to both the drain thereof on node  1410  and to the gate of an N-channel transistor  1414 . Transistor  1414  has the source/drain path thereof connected between the other side of transistor  1408  and V SS . Additionally, an offset transistor(s)  1416  of the P-channel type has the source/drain path thereof connected across the source/drain path of transistor  1408 , the gate thereof connected to V REF  and also to the gate of transistor  1408 . Transistor  1416  represents selectable transistors that are mask programmable to select a predetermined offset in the comparator. This offset at the input of the comparators aid in the supply independence. Without offset, the following would be true: With offset: 
 
 T   Period =2*(−τ*1 n (1 −V   Trip   /V   DD )+ T   Delay(comp) ) 
 
 T   Period =2*(−τ*1 n (1−ratio)+ T   Delay(comp) ) 
 
 V   Trip =ratio* V   DD  
 
 Without offset: 
 
 V   Trip   =V   Trip   +V   OS  
 
 T   period =2*(−τ*1 n (1−ratio− V   os   /V   DD )+ T   Delay(comp) ) 
 
         [0063]     From these equations, it can be seen that V DD  dependence has been added. Power supply dependence can be added or subtracted by varying the transistors  1416 , noting that there could be variable transistors across transistor  1406  also. This way, the offset can be made negative or positive. Again, this is a mask programmable system.  
         [0064]     Referring now to  FIG. 15 , there is illustrated a block diagram of two computer peripheral devices, a transmit device  1502  and a receive device  1504 , separated from each other and connected together through a serial communication line  1506 . Illustrated in  FIG. 15  is a unidirectional transmission path wherein information is transmitted from the transmitter  1502  over to the receiver  1504 . Associated with the transmitter is a transmit clock  1508  labeled T ref . The receiver  1504  has associated therewith a receive clock  1510  T clk . The data transmitted is illustrated in  FIG. 16  wherein there are a plurality of data bursts  1602  that occur over a timeline at different times although they may be repeated with a given periodicity. However, for bursty communications, it is just important to note that the communication actually disappears for a finite amount of time between bursts  1512 , such that the receive clock  1510  has a more difficult time capturing the clock information from the receive data.  
         [0065]     Referring now to  FIG. 17 , there is illustrated an overall block diagram for the system for recovering the clock from the bursty communication. The receive clock  1510 , as was described hereinabove, is a variable frequency clock and can have the frequency and phase thereof varied. In order to track data transitions and determine if the clock is out of lock, a first counter  1702  is provided, which first counter  1702  is referred to as a Slip Period Counter (SP counter). This counter  1702  is operable to receive as inputs the data transitions on a line  1704  and also the output of the clock circuit  1510  on a line  1706 . The receive clock  1510  is operated at a higher rate than the transmit clock, 4× in the present disclosure, to provide an oversampled condition. Therefore, there will be four cycles of the clock  1510  for each cycle of the transmit clock T ref . The Slip Period Counter  1702  is operable to initiate a count at a value of “0” upon the detection of the data transition. The counter  1702  will continue to count upward with the most significant bits with only the two least significant bits providing a modulo- 4  count, it being understood that this could be any modulus base, i.e., modulo- 8 , modulo- 16 , etc. Therefore, the two least significant bits will count from “0” through “3” and then count over again from “0.” The value of the two least significant bits is referred to as the phase slip value “PH.” In effect, these least two significant bits of the counter  1702  will provide four count values for each cycle of the transmit clock.  
         [0066]     As will be described hereinbelow, the slip period counter is reset on the occurrence of a phase slip and is then operable to use the PH value to determine when the next slip occurs. This will happen when a subsequent data transition is sampled and the value of the PH counter is not zero. This indicates that the received data interval is not an integer multiple of the PH clock period over four. For example, the counter  1702  is initiated upon a data transition in the received data. This will cause the count value to be incremented by the receive clock  1510 . The first data transition will occur at a PH clock value of “0.” If the transmit clock and receive clock are locked and there is no frequency error, and the data transition intervals are integer multiples of the PH clock period over four, then the data transition will always occur on a count value of “0” for the PH clock. However, if one of the clocks is drifting relative to the other of the transmit/receive clocks, then there will come a point in time where the value of the PH clock associated with the occurrence of a data transition may be either “1” for a transmit clock that is slow relative to the receive clock or a “3” value for a transmit clock that is running faster than the receive clock. The SP counter  1702 , upon determining that a data transition occurs on PH counter value of other than “0,” which constitutes a “slip,” will be noted and output to a processor  1710  and then the SP counter  1702  reset.  
         [0067]     A second counter, a counter  1708  is provided to count the number of receive clock cycles associated with a “bit period,” this referred to as a “BP” counter. The Bit Period is referred to as the number of receive clock cycles that occur between the two data transitions that occur between Mth data and the M- 1  data transition of the receive data, wherein the Mth data transition constitutes the data transition determined to be where the slip occurs and the SP counter  1702  was reset. The output of the BP counter  1708  is then latched in a latch  1712 , the output thereof provided to the processor  1710 . As will be described hereinbelow, the contents of the latch comprise the count for the last bit period for both the current slip period and the immediately previous slip period.  
         [0068]     In addition to determining the SP and BP count values, there will also be provided a master error (ME) block  1716 , which is utilized to calculate an error value. This utilizes both the data transition and the clock signal on lines  1704  and  1706 . This is provided to the processor  1710 .  
         [0069]     The processor  1710  will utilize the slip indication from the SP counter  1702 , in addition to the value thereof, the contents of the latch  1712  and the contents of the ME block  1716  in order to determine both the direction of an oscillator correction and also the magnitude of that correction required to effect a change in the receive clock  1510  to reduce the frequency error between the receive and transmit clocks. This is then output to an oscillator correction block  1720  that provides the correction control to the receive clock  1510 . This is to be contrasted with a standard phase lock loop that utilizes an iterative procedure wherein it calculates an error and then steps the oscillator in one direction or the other to again determine the error. The steps are typically constant steps. In the disclosed embodiment, the processor  1710  actually determines a value and a direction for the step. The correction block  1720  is comprised of a signed adder. This includes a summer  1721  that receives the error ε osc  on an input  1723 , the output of the summer  1721  input to a calibration register block  1730 . The output of the calibration register block  1730  is then input to the other input of the summer  1721  and provides the frequency input to the oscillator  1510 .  
         [0070]     In the clock recovery of the present disclosure, the period between incoming transitions is assumed to be an integer multiple of the period of the reference clock T ref , since a requirement of clock recovery is the ability to generate a reliable measure of a frequency error between the local oscillator and the reference or transmit oscillator. The present system utilizes the quantized nature of the incoming transition periods in order to generate a measure of the error. For example, if each of the incoming transitions is measured using the SP counter  1702  running at K times the local clock, then each transition should occur on the same modulo-K boundary of this SP counter  1702  when the receive and transmit clock are at the same frequency. However, as noted hereinabove, if the receive clock is slightly faster, the counter value modulo-K sampled at each incoming transition will tend to “walk” in the positive direction. Likewise, if the receive clock is slightly slower, the modulo-K sampled transition will tend to walk in the negative direction.  
         [0071]     Referring now to  FIG. 18 , there is illustrated a diagrammatic view of how the SP counter operates relative to the PH modulo-K counter. It can be seen that the SP counter is initiated at a value of “0” and then counts upward. The PH counter is also reset to zero at the same time as the SP counter, since this basically comprises the least two significant bits of the SP counter  1802 , and begins counting from “0” through “3.” This continually recycles, such that, for example, each multiple of 4 will result in a value of “0,” such as at the values “8” and “12.” 
         [0072]     Referring now to  FIG. 19 , there is illustrated a diagram of the SP count value from a value of “0” to a value of “SP.” If one transition is sampled at PH≠0 and a subsequent transition is sampled at PH=1, this is referred to as “phase slip,” which implies that the receive clock is faster than the reference clock. How much faster is not known, since even a slight mismatch in the clocks would eventually create this situation. In  FIG. 19 , the first transition is illustrated as occurring at the beginning of the SP count, this having been initiated by detecting a slip in a previous SP count value, as will be described in more detail hereinbelow. At a first count cycle  1902  for a count value of “0,” the transition could have been sampled anywhere within the clock cycle between the beginning of the count cycle to the next clock edge that will increment the counter. This is measured as a distance from the initial edge, an edge  1904  defined by the receive clock, to a transition edge  1906 , defined by the transmit clock T ref , which has a value relative to the receive clock, T clk , of a·T clk . The receive system will continue to sample the transitions until a phase slip is detected. This will occur at the last value in a slip period, SP, at a transition  1908 , again this transition determined by the transmit clock T ref . Transitions  1906  and  1908  are separated by an integer number of transmit clocks, N, giving an interval of N·T ref . This second transition will occur somewhere within a last count cycle  1910  of the SP counter  1702 . However, again, the position within the count cycle from one receive clock edge to the receive data transition is unknown. It is defined relative to the receive clock as a value of b·T clk . Knowing the time between phase slips and the value of the phase slip provides the frequency error, i.e., the change in phase over the change in time. However, it is noted that a phase slip pf “1” could represent jut greater than “0” to just less than “2” phase slips, due to the uncertainty a clk . The oscillator has a know maximum error and, therefore, if the transitions occur far enough apart in time such that the phase slip will be too large and exceed this error, then this measurement will be ignored. This condition can exist due to the bursty nature of the data and the potential for large dead times, resulting in the potential for too large a time between transitions. If not ignored, this could result in a frequency correction that would be of the wrong magnitude and/or sign.  
         [0073]     Referring now to  FIG. 20 , there is illustrated a more detailed diagrammatic view of the receive data transitions relative to the SP counter. The counter is reset at a value of zero at a first count cycle  2002  which is coincident with the occurrence of a receive data transition  2006  that is disposed a value of a T clk  from the leading edge of the count cycle  2002 . The transition  2006  is labeled T 1 . An nth transition  2004  labeled T n  is illustrated as occurring during the count cycle  2008  of the SP counter  2202  having a value of “41” with a PH value of “1,” indicating a phase slip of one SP clock. The transition  2004  is disposed a distance b·T clk  from the leading edge of the count cycle  2008 . If the clocks were locked, such that NT ref =NKT clk  with no relative phase shift, then this would result in a transition  2004 ′ for an nth transition T′ n  in a count cycle block  2010  with a value of “40” and a PH value of “0,” indicating no phase slip. The distance of the transition  2004 ′ from the edge of the count cycle block  2010  would be equal to a·T clk , such that the time between transition  2006  and transition  2004 ′ would be NKT ref . However, since there is a phase slip of the clock, then the value from the leading edge of SP clock  2002  would be N clkf +(b−a+1)·T clk , resulting in an error  2012 . This error can be utilized to adjust the clock, if both the sign of that error is known and the magnitude of that error is known. The reason for this error is that each incoming transition in the SP counter includes the uncertainty factors (a &amp; b) which range from zero to one. The measured error between the clocks can be derived as:  
             1.   ⁢           ⁢     
     ⁢           ⁢         aT   clk     +     N   ·     T   ref         =       SP   ·     T   clk       +     bT   clk                     2.   ⁢           ⁢     
     ⁢           ⁢         T   clk       T   ref       =     N     SP   +   b   -   a                     3.   ⁢           ⁢     
     ⁢           ⁢           T   ref     ⁡     (     1   +   ɛ     )         KT   ref       =     N     SP   +   b   -   a                     4.   ⁢           ⁢     
     ⁢     ɛ   =         KN     SP   +   b   -   a       -   1     =       KN   -   SP   -   b   +   a       SP   +   b   -   a                       5.   ⁢           ⁢     
     ⁢           ⁢     ɛ   =       PH   -   b   +   a       SP   +   b   -   a                     6.   ⁢           ⁢     
     ⁢           ⁢         PH   -   1       SP   +   1       &lt;   ɛ   &lt;       PH   +   1       SP   -   1                   
 
         [0074]     Note that, in these equations, PH is treated as a 2&#39;s complement value, i.e., 3=−1, etc. In equation 6, it is noted that for small absolute values of PH (−1,1), that only the sign information is known and the error magnitude is still unknown. The BP counter  1708  is utilized to help recover the magnitude information. In addition, using the BP counter and the SP counter, it is known that there are either two positive phase changes in a row or two negative phase changes. If there is a positive and then a negative or just the reverse, then the error is ignored.  
         [0075]     Referring now to  FIG. 21 , there is illustrated a diagrammatic view of the operation of the SP counter  1702  and the BP counter  1708 . The value for the BP counter comprises the last Bit Period that occurred prior to and up to the resetting of the SP counter  1702 . For illustration purposes, there are illustrated two Slip Periods, a first Slip Period having a value from zero to SP 1  and a second Slip Period having a value from zero to SP 2 . The first Slip Period will have its final count value SP 1  occur upon a transition  2102  and the data that occurs for a PH value that is other than “0” (not shown). This will occur in a count cycle  2104 . At this time, the SP counter  1702  is reset to a value of “0,” such that the count cycle  2104  constitutes the first count cycle of the second Slip Period and the value of PH therefore will be set to “0.” This will continue to count up for an integral multiple of the transmit clock until a transition  2106  occurs that is associated with the PH value of other than “0” in a count cycle  2108 . Again, the first transition  2102  occurs a distance of a·T clk  from the leading edge of the count cycle  2104  and the transition  2106  occurs at a distance b·T clk  from the leading edge of the count cycle  2108 .  
         [0076]     With respect to the BP counter  1708 , the transition  2102  that resulted in resetting of the SP counter  1702  for the first Slip Period occurred an integral multiple of the T ref  clock from a prior transition  2110 . This prior transition  2110  is the immediately preceding transition to data transition  2102 . However, it should be understood that this data transition could occur at any multiple of the transmit clock, depending upon the communication protocol utilized. The BP counter  1708  is reset on each transition of the data, such as transition  2110  for the last Bit Period in the first Slip Period in a count cycle  2112 , a distance of c·T clk  from the leading edge of the count cycle  2110 . The BP counter  1708  will be incremented up to a value BP in a count cycle  2114  that corresponds to the SP count cycle  2104 . This will be a distance of a·T clk  from the leading edge thereof. When the transition  2102  occurs and a slip is detected, the BP count value is stored and this value is utilized to calculate the clock error.  
         [0077]     Similar to the previous case, for this case, the clock error can be calculated as follows:  
             7.   ⁢           ⁢     
     ⁢         cT   clk     +     M   ·     T   ref       +     N   ·     T   ref         =       BP   ·     T   clk       +       SP   2     ·     T   clk       +     bT   clk                     8.   ⁢           ⁢     
     ⁢         T   clk       T   ref       =       M   +   N       BP   +     SP   2     +   b   -   c                     9.   ⁢     
     ⁢           ⁢           T   ref     ⁡     (     1   +   ɛ     )         KT   ref       =       M   +   N       BP   +     SP   2     +   b   -   c                     10.   ⁢     
     ⁢           ⁢     ɛ   =       KM   +   KN   -   BP   -     SP   2     -   b   +   c       BP   +     SP   2     +   b   -   c                     11.   ⁢           ⁢     
     ⁢           ⁢     ɛ   =         PH   1     +     PH   2     -   b   +   c       BP   +     SP   2     +   b   -   c                   
 
         [0078]     It can be shown that (KN−SP 2 )=PH 2  and (KM−BP)=PH 1 , such that:  
       12.   ⁢               
               ⁢                  PH   1     +     PH   2            -   1       BP   +     SP   2     +   1       &lt;        ɛ        &lt;                PH   1     +     PH   2            +   1       BP   +     SP   2     -   1             
 
 PH 1  and PH 2  are those values associated with SP 1  and SP 2 , respectively. 
 
         [0079]     Equation 12 defines the bounds of the error as to the magnitude thereof. The sign has already been determined and this equation, for any non-zero value of PH 1  and PH 2 , results in a non-zero lower limit on a clock error that can be calculated. Therefore, a frequency correction of up to twice this lower limit can be made to the local clock without risking an increase in the resulting absolute error. The reason for this is that a determined error of, for example, +1.5% on the lower limit could be adjusted by up to 3% in the opposite direction that would result in the clock then having a −1.5% error, which would result in no worse error. The correction factor that is generated is as follows:  
     13.     
         ɛ   osc     =       SIGN   ⁡     (       PH   1     +     PH   2       )       ⁢     (                PH   1     +     PH   2            -   1       BP   +     SP   2     +   1       )     ⁢     (     1   +   o     )           
 
 where o is the maximum allowed overshoot and the sign of the correction is the same as that of the PH values. This equation 13 therefore provides both the sign and the magnitude for a given overshoot factor “o.”
 
         [0080]     The derivations above use the relationship that (KN−SP 2 )=PH 2 . However, since we sample the value of PH 2  on incoming transitions, aliasing can occur thus making this relationship invalid. The rate of change of PH 2  relative to edges on T ref  is equivalent to the clock error over KT clk . Thus the Nyquist Limit requires that:  
     14.     
         1     BP   ·     T   clk         &gt;     2   ⁢     ɛ     KT   clk             
 
         [0081]     Where BP is the number of T clk  periods between incoming transitions. In general, the maximum value of BP which avoids aliasing is given by:  
     15.     
         BP   max     &lt;     K     2   ⁢   ɛ           
 
         [0082]     Transitions which result in a BP count longer than this amount should be ignored to avoid errors. Notice that in order to know when to reject incoming transitions, an upper limit on the clock error must be known. Although a worst-case error could be used based on the starting oscillator tolerance, this would tend to decrease the bandwidth of the tracking loop by rejecting more transitions, thus increasing the time required to reduce the clock error to within acceptable limits. An alternative is to maintain an upper limit for the clock error calculated from incoming transitions (named ME). This limit can then be used to calculate the value of BP max . This upper limit can be initialized to the worst-case oscillator error. As incoming transitions are observed, ME can be reduced, thus increasing BP max  and allowing larger gaps between useful transitions. This is of particular importance when dealing with bursty communications systems, such as USB. Calculation of ME will be described hereinbelow.  
         [0083]     Although the above equation is correct in general, further considerations can be used to refine the BP max  calculations. For example, since the SP and hence PH values are reset to “0” after each incoming transition, this effectively locks the phase of T ref  with PH. Therefore, for a given sampled value of PH, a specific value for BP max  can be calculated which prevents aliasing for that specific PH value. This is accomplished by considering how the phase error between T ref  and PH accumulates over time. This accumulated error must be limited to prevent the sampled value of PH from wrapping. Although in general this implies the phase error must remain less than 180 degrees, since T ref  and PH are phase-locked on each transition, the allowed phase error can exceed 180 degrees in some cases. The error accumulated over M T ref  periods is given by: 
 
Δ T=aT   clk   +M ( KT   clk   −T   ref )   16. 
 
 where a is the initial phase error. To avoid aliasing, this error must not exceed the cardinal distance between K and PH (named D). Therefore: 
 
 aT   clk +M( KT   clk   −T   ref )&lt; DT   clk    17. 
 
 where D is given by:  
     18.     
       D   =     {           K   -        PH              if              PH        &lt;     K   2                   2   ⁢   K     -        PH              if           PH        =     K   2                   
 
         [0084]     Using: 
 
 T   ref   =KT   clk /(1+ε)   19. 
 
 and solving for M gives:  
     20.     
       M   &lt;         (     D   -   a     )     K     ⁢       (     1   +   ɛ     )     ɛ           
 
         [0085]     Using the relationship between the clocks and a worst-case value of “1” for a, this can be written in terms of BP max T clk  periods as:  
               BP   max     &lt;       D   -   1     ɛ           21.           
 
         [0086]     Note that this value of BP max  is the largest value allowed which rejects aliasing transitions. However, due to the quantized nature of PH values, a lower limit on BP max  can be calculated which guarantees no valid transition for a given sampled value of PH will be rejected. This is obtained by requiring the accumulated error always exceed the cardinal distance between PH and 0 plus 1, written as: 
 
 aT   clk   +M ( KT   clk   −T   ref )&gt;( PH +1)· T   clk    22. 
 
         [0087]     This can be solved in terms as BP max  using a value of 0 for a as:  
               BP   max     &gt;            PH        +   1     ɛ           23.           
 
         [0088]     For a given implementation of this clock recovery algorithm, using any value of BP max  which satisfies both limits will prevent aliasing without rejecting acceptable incoming transitions. The actual value used can be chosen within these limits in a manner which reduces hardware complexity.  
         [0089]     As discussed above, a measure of the maximum clock error is required for proper rejection of aliasing transitions. Starting with Eqn. 6, the maximum absolute error can be written as:  
                    ɛ   max          &lt;     {                  PH   -   1            SP   +   1               if   ⁢           ⁢   PH     &lt;   0                 PH   +   1       SP   -   1               if   ⁢           ⁢   PH     ≥   0                   24.           
 
         [0090]     In order to simplify the hardware, this can approximated without loss of generality as:  
                    ɛ   max          &lt;            PH        +   1       SP   -   1             25.           
 
         [0091]     The maximum error register (named ME) can be initialized with the worst-case initial oscillator error. On each valid incoming transition, the current value of SP can be used to calculate a new absolute value Of ζ max . If this new term is less than ME, ME can be reduced to this new value. BP max  can then be calculated using ζ=ME. This measure of the maximum error improves over time, thus allowing larger gaps between incoming transitions before aliasing can occur. Note that if a worst-case oscillator drift over time is known, the value of ME can be increased by a correction factor at regular intervals in order to account for this drift.  
         [0092]     Referring now to  FIG. 22 , there is illustrated a block diagram illustrating the counters and the implementation of the various equations noted hereinabove that are required to implement the clock recovery algorithm of the present disclosure. A general control block  2202  is provided that is operable to receive the data transition on a transition line  2204  and generate on an output therefrom an oscillator correction request on a line  2206 . The BP counter  1708  is illustrated as having an accumulating register  2208  that receives as an input the output of a two input multiplexer  2210 . The multiplexer  2210  is controlled by the data transition to, upon the occurrence of a data transition, select a “1” for a reset operation and, in the absence of a data transition, a feedback increment loop is selected which, upon each clock cycle of the receive clock, will increment the value in the BP latch  2208 . The operation is such that the value stored in the BP latch  2208  will be loaded into one input of a multiplexer  2212  on the “1” input which multiplexer  2212  is controlled by a slip detection output  2214  from the control block  2202  that will cause the value of the BP latch  2208  to be loaded into a BP save  latch  2218 . In the absence of the Slip Signal, the output of the latch  2218  is fed back to the “0” input of the multiplexer  2212 , such that it is continually maintained in the latch  2218 .  
         [0093]     The SP counter  1702  is realized with an SP latch  2220  that receives as the input thereof the output of the two input multiplexer  2222 , the “1” input connected to a fixed reset value of “1,” the multiplexer providing a reset when the Slip Signal is indicated on line  2214 . In the absence of the Slip Signal, the output of latch  2220  is fed to an increment block  2224  that is fed back to the “0” input of multiplexer  2222  such that, for each clock cycle, the value of the SP counter  2220  is incremented. Upon the occurrence of a Slip Signal, the output of the latch  2220  is input to an algorithm block  2226  to execute equation 13 to determine the oscillator period correction factor, as well as the value out of the BP save  register  2218 . These two values are utilized to perform this operation.  
         [0094]     The output of the SP latch  2220  is also input back to the control block  2202  to determine the PH values therefor. These PH values are what are utilized to determine if a slip has occurred. This SP output value is also input to a maximum error calculation block  2228  to calculate the maximum error value, the output of block  2228  input to the A-input of block  2234 . This is input to the “1” input of a two input multiplexer  2230 , the output of which feeds an ME register  2232  that provides the ME output. This is input back to the “0” input of the multiplexer  2230 . The output of the calculation block  2228  is also input to a comparator block  2234  having A and B inputs which B input is connected to the output of the ME latch  2232 , it being understood that the initial value of the ME latch  2232  is the maximum error value that is predetermined for the system. The block  2234  is operable to determine if the value of A is less than the value of B. If so, this is logically ANDed with the transition input  22  so, if both the condition that A is less than B and the data transition occurs at the same time, then this causes the value calculated in block  2228  to be loaded into the ME block  2232 . The output of ME latch  2232  is then input to a calculation block  2236  which also receives the output of the SP latch  2224  calculating the anti-aliasing value and determining the value of BP max . This is input to the A-input of a comparator block  2238 , the B-input connected to the output of the BP latch  2208 . If the A-input is determined to be less than the B-input, this indicates that there is an alias condition, which is indicated back to the control block  2202  for the purpose of possibly determining that the slip indication has not actually occurred and it will be ignored.  
         [0095]     Referring now to  FIG. 23 , there is illustrated a simplified flowchart depicting the basic control steps. The program is initiated at a Start block  2302  and then proceeds to a function block  2304  to set the value of ME to a maximum initial error and then to a function block  2306  to wait for an incoming transition. When the incoming transition occurs, the BP counter is reset, as indicated by function block  2308 . The program then flows to a function block  2310  to wait for the next transition. When the next transition occurs, both the SP counter and the BP counter are sampled, as indicated by a function block  2312 . The program then flows to a decision block  2314  to determine if the value of BP is less than the value of BP max . If so, the program will flow along a “Y” path to a function block  2316  to update the value of ME and then to a decision block  2318  to determine if the value of PH is equal to zero. If not, this indicates a slip and then the program proceeds to a decision block  2320 . If yes, then the program will flow back to the input of the function block  2308 . Similarly, if anti-aliasing is required and BP is determined to be not less than BP max  at decision block  2314 , then the program will flow to a function block  2322  from decision block  2314  to reset the SP counter and then back to the input of function block  2308 . However, when the PH value is determined not to be zero, i.e., a slip condition, then the program flows to decision block  2320 , and a decision is made as to whether a current value of PH and the previous value of PH have the same sign. If not, the program will flow to function block  2322  in order to reset the SP counter and if so, the program will flow along the “Y” path to a function block  2328  to correct a local oscillator by the minimum determined error.  
         [0096]     Implementation of the algorithm described above is largely an exercise in managing quantization and finite-register effects. For example, the oscillator correction factor must be specified as an integer multiple of the oscillator&#39;s inherent period resolution. Also, since the logic must never increase the relative error, the oscillator&#39;s maximum possible unit step size must be used in the calculations. This value is defined in the RTL code as:  
             MAXOSCSTEP   =     1     ɛ   step             26.           
 
         [0097]     The initial worst-case oscillator error (ε init ) is defined as a multiple of ε step , i.e.:  
             MAXINITERROR   =       ceil   ⁡     (       ɛ   init       ɛ   step       )       =     ceil   ⁡     (     MAXOSCSTEP   ·     ɛ   init       )               27.           
 
         [0098]     The maximum allowed overshoot is defined as a multiple of 25%, i.e.: 
 
OVERSHOOT=4o   28. 
 
         [0099]     Finally, the Maximum Error (ME) is maintained as an integer multiple of ε ME (a fixed fraction of ε step ), defined by ERRORSTEP as:  
             ERRORSTEP   =         ɛ   step       ɛ   ME       =     1     MAXOSCSTEP   ·     ɛ   ME                 29.           
 
         [0100]     The width of the various counters is limited in general by the maximum values for which a non-aliasing calculation would be performed.  
         [0101]     The remaining implementation issue is how to avoid the division inherent in most of the equations described herein. For the value of K=4 used in the implementation, most of the equations only have eight or fewer unique values. This allows the divisions to be precomputed. For example, consider the calculation of the oscillator correction factor:  
                    ɛ   osc          =       (         PH   1     +     PH   2     -   1       BP   +     SP   2     +   1       )     ⁢     (     1   +   o     )             30.           
 
         [0102]     This can written as:  
               BP   +     SP   2       =         (         PH   1     +     PH   2     -   1            ɛ   osc            )     ⁢     (     1   +   o     )       -   1           31.           
 
         [0103]     Now, for K=4 the only interesting cases correspond to (PH 1 +PH 2 )=±2 (the case of (PH 1 +PH 2 )=±3 is approximated as ±2 to simplify the hardware). Therefore, the equation can be written as:  
               BP   +     SP   2       =       (       1   +   o            ɛ   osc            )     -   1.           32.           
 
         [0104]     The value of ε osc  can be quantized to 3 bits since 2 3 ε step &gt;ε init . Therefore, the oscillator correction factor can be calculated as:  
                    ɛ   osc          =     {         7             if   ⁢           ⁢   BP     +     SP   s       &lt;       (       1   +   o       7   ⁢     ɛ   step         )     -     1   ⁢           ⁢   else                 6             if   ⁢           ⁢   BP     +     SP   2       &lt;       (       1   +   o       6   ⁢     ɛ   step         )     -     1   ⁢           ⁢   else                 5             if   ⁢           ⁢   BP     +     SP   2       &lt;       (       1   +   o       5   ⁢     ɛ   step         )     -     1   ⁢           ⁢   else                 4             if   ⁢           ⁢   BP     +     SP   2       &lt;       (       1   +   o       4   ⁢     ɛ   step         )     -     1   ⁢           ⁢   else                 3             if   ⁢           ⁢   BP     +     SP   s       &lt;       (       1   +   o       3   ⁢     ɛ   step         )     -     1   ⁢           ⁢   else                 2             if   ⁢           ⁢   BP     +     SP   s       &lt;       (       1   +   o       2   ⁢     ɛ   step         )     -     1   ⁢           ⁢   else                 1             if   ⁢           ⁢   BP     +     SP   s       &lt;       (       1   +   o       1   ⁢     ɛ   step         )     -     1   ⁢           ⁢   else                 0                             33.           
 
         [0105]     This corresponds to a set of seven comparators plus some encoding logic in hardware.  
         [0106]     Referring now to  FIG. 24 , there is illustrated a diagrammatic view of one instantiation of the precision oscillator. In the oscillator implemented on the integrated circuit, a programmable internal clock generator  2402  is provided that is controlled by a register  2406  and a register  2408 . The output of the internal clock generator is input to a divide circuit  2410 , which is also controlled by the register  2408 , the output thereof being input to one input of a multiplexer  2410 . This multiplexer  2410  is controlled by the register  2408 . Register  2410  outputs the system clock (SYSCLK), which is input to the baud rate generator  302 . In addition to an internal clock generator, there is also a provision for an external crystal controlled oscillator. A crystal controlled internal or on-chip oscillator  2412  is provided that is interfaced through an input  2414  to terminals  2416  and  2418  to an external crystal  2416 . The output of the oscillator  2412  is input to one input of the multiplexer  2410 . Additionally, an external clock is provided on a terminal  2420  that is also input to one input of the multiplexer  2410 . The crystal controlled oscillator  2412  is controlled by a register  2422 .  
         [0107]     The internal oscillator is provided such that it will be the default system clock after a system reset. The internal oscillator period can be programmed with the register  2406  by the following equation:  
         Δ   ⁢           ⁢   T     ≅     0.0025   ×     1     f   BASE       ×   Δ   ⁢           ⁢   OSCICL         
 
 wherein f BASE  is a frequency of the internal oscillator followed by a reset, ΔT is the change in internal oscillator, and ΔOSCICL is a change to the value held in the register  2406 . Typically, the register  2406  will be factory calibrated to a defined frequency such as, in one example, 12.0 MHz. 
 
         [0108]     Referring now to  FIG. 25 , there is illustrated a table for register  2406  wherein it can be seen that bits  6 - 0  are associated with the calibration register of the oscillator and its value can be changed internally.  FIG. 26  illustrates the control register  2408  illustrating the controls provided therefor.  
         [0109]     Although the preferred embodiment has been described in detail, it should be understood that various changes, substitutions and alterations can be made therein without departing from the spirit and scope of the invention as defined by the appended claims.