Abstract:
A bandgap surface for use in a waveguide transition module. The surface may be constructed with a ground plane; a capacitive layer including a plurality of capacitive elements arranged in a polar configuration within a common plane, and spaced apart from said ground plane; and a plurality of electrically conductive vias formed within a dielectric layer of material that conductively couple said ground plane to each of said capacitive elements.

Description:
FIELD 
     The present disclosure relates to impedance surfaces, and more particularly to more compactly constructed impedance surfaces and waveguide transition modules making use of such impedance surfaces. 
     BACKGROUND 
     The statements in this section merely provide background information related to the present disclosure and may not constitute prior art. 
     Current coaxial to waveguide transition modules require a quarter wavelength back short between the conductive elements and a ground plane. For example, for operations at 15 GHz using a construction material with a relative dielectric of 2.2 and a circular waveguide diameter of 0.340 inch, the quarter wavelength is approximately 0.350 inch thick. Put differently, the quarter wavelength back short will have a height of about 0.350 inch. When this transition module is fabricated into a printed wiring board that also contains clock, data, voltage, ground planes and multiple stripline layers (as required for multiple polarizations or frequencies), the final stackup height of the overall assembly can easily exceed about 0.450 inch in height. Such a height is undesirable for a number of applications, and particularly phased array antenna applications, where the compactness of the module is of the utmost importance. 
     Existing solutions typically require construction of two separate wiring boards, one containing the quarter wavelength back short and one containing the additional circuits. The separate wiring boards are then joined using an epoxy bond process. As will be appreciated, the requirement of having to join the two assemblies with epoxy adds to the manufacturing complexity and cost of producing a waveguide transition module. The resulting module may also be undesirably thick. 
     SUMMARY OF THE INVENTION 
     In one aspect the present disclosure relates to a bandgap surface for use in a transition module. The surface may comprise: a ground plane; a capacitive layer including a plurality of capacitive elements arranged in a polar configuration within a common plane, and spaced apart from the ground plane; and a plurality of electrically conductive vias formed within a dielectric layer of material that conductively couple the ground plane to each of the capacitive elements. 
     In another aspect the present disclosure relates to a dual frequency impedance surface for use in a transition module. The impedance surface may comprise: a first impedance surface forming a plane, and adapted to resonate at a first frequency; and a second impedance surface placed within a void in the first impedance surface so as to be disposed within the plane and circumscribed by the first impedance surface, the second impedance surface being adapted to resonate at a second frequency different than the first frequency. 
     In another aspect the present disclosure relates to a waveguide for transitioning between transverse electromagnetic (TEM) wave and transverse electric (TE) wave modes. The waveguide may comprise: a tubular waveguide structure for conducting a TE wave; a polar geometry impedance surface disposed within the tubular waveguide structure generally normal to a longitudinal axis of the tubular waveguide structure; a pair of coaxial transmission line sections extending orthogonal to the polar geometry impedance surface, and terminating adjacent to the polar geometry surface, for conducting a TEM mode wave; and first and second conductive elements arranged generally at a right angle to one another, and positioned over the polar geometry impedance surface adjacent the coaxial transmission line segments, for interfacing signals passing between the coaxial transmission line sections and the polar geometry surface such that the signals transition between the TEM and TE wave modes. 
     In still another aspect the present disclosure relates to a waveguide for transitioning between two transverse electromagnetic (TEM) wave modes and two transverse electric (TE) wave modes. The waveguide may comprise: a tubular waveguide structure for conducting a TE wave; a polar geometry impedance surface for resonating at a first frequency, and being disposed within said tubular waveguide structure generally normal to a longitudinal axis of the tubular waveguide structure; a first pair of coaxial transmission line sections terminating adjacent the polar geometry impedance surface, for conducting a first TEM mode wave between the first pair of coaxial transmission line sections and the polar geometry impedance surface; and first and second conductive elements arranged generally at a right angle to one another, and positioned over the polar geometry impedance surface and coupled to the first pair of coaxial transmission line segments, for interfacing signals passing between the first pair of coaxial transmission line sections and the polar geometry impedance surface such that the signals transition between the TEM and TE wave modes; a second impedance surface for resonating at a second frequency, and being disposed adjacent the polar geometry impedance surface; and third and fourth conductive elements arranged generally at right angles to one another, and arranged over the second impedance surface adjacent connected to conductive traces, for interfacing signals passing between the second impedance surface and the ground plane such that the signals transition between the TEM and TE wave modes. 
     In still another aspect the present disclosure relates to a method for forming a bandgap surface as a multilayer board assembly. The method may comprise: forming a ground plane as a first layer; forming a capacitive layer as a second layer using a plurality of capacitive elements arranged in a polar configuration within a common plane; disposing the second layer adjacent the first layer; and forming the first layer to include a plurality of electrically conductive vias that electrically couple the ground plane to each of the capacitive elements. 
     Further areas of applicability will become apparent from the description provided herein. It should be understood that the description and specific examples are intended for purposes of illustration only and are not intended to limit the scope of the present disclosure. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The drawings described herein are for illustration purposes only and are not intended to limit the scope of the present disclosure in any way. 
         FIG. 1  is a perspective view of an assembly forming a bandgap surface in accordance with one embodiment of the present disclosure; 
         FIG. 2  is a plan view of the assembly of  FIG. 1 ; 
         FIG. 3  is a plan view of another embodiment of the present disclosure that makes use of an additional layer of capacitive elements; 
         FIG. 4  is a side view of the assembly of  FIG. 3 ; 
         FIG. 5  is a perspective view of the assembly of  FIG. 3  incorporated into a circular waveguide; 
         FIG. 6  is a graph illustrating the reflection phase of a polar geometry, high impedance ground plane, and more particularly the reflection performance (in degrees) of the waveguide of  FIG. 5  at various frequencies between 28 GHz and 34 GHz, relative to a normal ground plane reflection phase; 
         FIG. 7  is a plan view of another embodiment of the present disclosure that forms a first waveguide within a second waveguide; 
         FIG. 8  is a perspective view of a waveguide transition module incorporating the assembly of  FIG. 3  for transitioning a TEM wave mode to a TE wave mode, and vice versa; 
         FIG. 9  is a graph entitled “Single Frequency Module with Polar Geometry EBG Surface”, which illustrates the insertion loss, the return loss and isolation of signals (all represented by “Y 1 ” in dBs relative to frequency, in GHz) passing through the assembly of  FIG. 8 ; 
         FIG. 10  is a perspective illustration of a waveguide transition module that transitions two coaxial (TEM) modes to two waveguide (TE) modes, and vice versa; 
         FIG. 11  is an enlarged view of a center portion of the module of  FIG. 10 ; 
         FIG. 12  is a side view of the module of  FIG. 11 ; 
         FIG. 13  is a graph showing the low frequency performance, in terms of insertion loss, isolation and return loss of the dual frequency module (in dBs relative to frequency, in GHz) of  FIGS. 11 and 12 ; and 
         FIG. 14  is a side view illustrating exemplary layers of a printed wiring board forming an assembly stackup that makes use of the polar geometry bandgap surface of  FIG. 1 . 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The following description is merely exemplary in nature and is not intended to limit the present disclosure, application, or uses. It will also be appreciated that like numbers used in the written specification correspond to similar components in the drawings. Thus component  100 , for example, if shown in  FIG. 7 , relates to the same component shown and discussed in  FIG. 3 , even though no additional explicit description of component  100  may be provided in the written discussion of  FIG. 7 . 
     Referring to  FIGS. 1 and 2 , there is shown one embodiment of an assembly  10  or “stackup” that forms a polar geometry bandgap surface. The assembly  10  makes use of a ground plane  12  formed on one surface of a dielectric layer  14 , a plurality of capacitive elements or plates  16   a  and  16   b  (referred to collectively as “capacitive elements or plates  16 ”) arranged on an opposite surface of the dielectric layer  14 , and a plurality of electrically conductive or inductive vias  18   a  and  18   b  (collectively referred to as “conductive vias  18 ”) formed within the dielectric layer  14  to electrically couple the ground plane  12  to each of the capacitive elements  16   a  and  16   b . The dielectric layer  14  may comprise Duroid 5880 or any other suitable dielectric and typically will be in the range of about 20-30 mils (0.508-0.762 mm) thick. The electrical vias  18  will typically have a diameter of about 0.005 inch (0.127 mm) and be manufactured from copper, gold or another suitable conductor. The vias  18  are arranged to extend orthogonal to the capacitive elements  16 . 
     The assembly  10  has the capacitive plates  16  arranged in a polar configuration with the central circular element  16   a  surrounded by the arcuately shaped capacitive elements  16   b . This is in contrast to previously developed bandgap surfaces that rely on an X-Y grid of capacitive elements. The assembly  10  forms an electromagnetic bandgap (“EBG”) surface that does not support an electric current and reflects a tangential electric (“E”) field with a phase shift of zero degrees (i.e., an open circuit). In contrast, a normal ground plane does support an electric current and reflects a tangential E field with a phase shift of 180 degrees. Any induced current on a conductive element placed adjacent to a normal ground will be cancelled due to the 180 degree phase shift while the induced current on a conductive element placed adjacent to an EBG surface will not. In previously developed systems using a normal ground plane, the conductive elements will be found a quarter wavelength above the ground plane where the total field path length will measure 1 wavelength. It can be appreciated, then, that an EBG ground surface will allow significantly more compact construction when paired with conductive elements. 
     Referring to  FIGS. 3 and 4 , another assembly  100  ( FIG. 3 ) is shown in accordance with another embodiment of the present disclosure. This embodiment makes use of the capacitive plates  16   a  and  16   b , as well as conductive vias  18   a  and  18   b . The assembly  100  is identical in construction to the assembly  10  but also includes a second layer (i.e., second plane) of capacitive elements  120   a  and  120   b  (referred to collectively as “capacitive elements  120 ”) formed on a dielectric layer  122  ( FIG. 4 ), and arranged in a polar configuration. The dielectric layer  122  is positioned on the dielectric layer  14 . The second layer of capacitive elements  120  operates to lower the tuned frequency of the assembly  100 . This is because the resonant frequency is determined by 1/sLC. 
     Referring to  FIG. 5  an apparatus  200  is shown that includes the apparatus  100  disposed within a circular waveguide structure  202 . The waveguide structure  202  will typically be filled with a suitable dielectric material, such as R03006, with the axial center of the apparatus  200  disposed in alignment with the axial center of the waveguide structure  202 . The apparatus  200  also includes components  16   a ,  16   b ,  18   b ,  120   a  and  120   b  in the manner described in connection with  FIGS. 1-4 . 
       FIG. 6  illustrates the reflection phase performance of the apparatus  100  of  FIG. 1 , with the graph of  FIGS. 6  showing reflection Y 1  in degrees, relative to the frequency in GHz. In this example the apparatus  100  is tuned for a resonant frequency of 31 GHz. It will be noted that the reflection Y 1  at the EBG surface (i.e., components  16   a  and  16   b ) of the apparatus  100  (in degrees) is approximately zero degrees at 31 GHz. Thus, the apparatus  100  forms a highly effective reflective band gap surface. The apparatus  100 , whose performance is illustrated in  FIG. 6 , includes components  14 ,  16   a ,  16   b ,  18   a ,  18   b ,  120 ,  120   a ,  120   b  and  122  as discussed in connection with  FIGS. 1-4 . 
     Referring to  FIG. 7  an apparatus  300  in accordance with another embodiment of the present disclosure is illustrated. The apparatus  300  effectively combines two bandgap surfaces that may be tuned to different frequencies. This is accomplished by incorporating the apparatus  100  within a well known Sievenpiper bandgap surface  302 , although a second bandgap surface of a polar configuration may also be used. The apparatus  100  is preferably arranged at the axial center of the Sievenpiper bandgap surface  302 . The Sivenpiper bandgap surface  302  as shown makes use of a plurality of capacitive elements  302   a  arranged in an hexagonal pattern. Other popular embodiments of the Sivenpiper bandgap surface  302  make use of rectangular capacitive elements arranged in an X-Y grid. Each capacitive element is coupled to its own electrically conductive via  303 . The apparatus  100  in this example forms a high impedance ground plane for high band operation, while the Sievenpiper bandgap surface  302  forms a high impedance ground plane for low band operation. The precise frequency band for the apparatus  100  and the Sievenpiper bandgap surface  302  is selected in accordance with the needs of a particular application. For example, the apparatus  100  may be tuned to a resonant frequency of 31 GHz while the Sievenpiper bandgap surface may be tuned to a lower frequency of 14.5 GHz. 
     Referring to  FIG. 8 , a waveguide transition module  400  is shown in accordance with another embodiment of the present disclosure. The module  400  includes the apparatus  100  positioned within a circular waveguide structure  402  filled with a dielectric material forming a plug  403 . Conductive elements  404 , (each having a first element  404   a  and a second element  404   b ) disposed on an additional layer of dielectric material that resides on the apparatus  100 , and the conductive elements  404   a  and  404   b  are arranged orthogonal to one another. The orthogonal elements  404   a  and  404   b  used in this embodiment allow the generation of waveguide field vectors in linear or circular polarizations. It will be understood that a single conductive element may be used in place of the dual orthogonal elements  404   a ,  404   b  if only a linearly polarized field vector is required. A pair of coaxial transmission lines  406  replaces the pair of conductive vias  18   b . The coaxial transmission lines  406  are also aligned with the conductive elements  404   a ,  404   b  and terminate such that the center conductor of each coaxial transmission line  406  contacts a respective one of the conductive elements  404   a ,  404   b . Thus, it will be appreciated that the apparatus  100  is modified to accommodate the formation of the coaxial transmission lines  406  so that the lines extend through the dielectric material  14  of the apparatus  100 . The waveguide transition module  400  operates to transition a coaxial mode (TEM) signal to a waveguide mode (TE) signal, or vice versa, depending on the direction of flow of the signal passing through the module. 
       FIG. 9  shows a graph that illustrates the insertion loss, return loss and isolation of the waveguide transition module  400  of  FIG. 8 , with the losses Y 1  (in dBs) plotted relative to frequency, in GHz. The insertion loss is ideally as low as possible while the return loss is ideally as high as possible. The isolation is ideally maximized at the resonant frequency. In this example, with the module  400  tuned for a resonant frequency of 32 GHz, the input signal is split evenly between different ones of the conductive element pairs  404   a / 404   b . The insertion loss between the conductive elements is virtually identical and is minimized at about 1 dB at the resonant frequency. The insertion loss shown in  FIG. 9  of 4 dB includes the 3 dB from the signal split. Conversely, the return loss of the conductive element pairs  404   a / 404   b  is ideally maximized at the resonant frequency.  FIG. 9  illustrates the return loss experienced by the conductive element pairs  404   a / 404   b  as being similar and maximized at the resonant frequency. The isolation of the conductive element pairs  404   a / 404   b  is also maximized at the resonant frequency. 
     Referring to  FIGS. 10 ,  11  and  12 , a dual band waveguide transition module  500  in accordance with another embodiment of the present disclosure is shown, which includes apparatus  100 . The module  500  makes use of dual frequency impedance surface apparatus  300  with capacitive element  302   a  ( FIG. 11 ), and further with the inclusion of a first pair of conductive elements  502  ( FIGS. 10 and 11 ) formed on a surface of a dielectric  504  ( FIGS. 11 and 12 ), and a dielectrically loaded circular waveguide  505  located above the dielectric surface  504 . The waveguide  505  has an axial center disposed in alignment with the axial center of the apparatus  300 . A second pair of conductive elements  506  ( FIGS. 10 and 11 ) is formed on a surface of dielectric  510  and a dielectrically loaded circular waveguide  507  ( FIGS. 11 and 12 ) is located above the dielectric surface  504 , itself located above dielectric surface  510 , with the axial center of the waveguide  507  disposed in alignment with the axial center of the apparatus  300 . Certain of the electrically conductive vias  303  are also visible in  FIG. 12 . The waveguide  505  located on dielectric surface  504  is centered within the waveguide  507  also located on dielectric surface  504 . In this example the second pair of conductive elements  506  are located 0.004″ below the starting point of waveguide  507  with the resulting gap not effecting electrical performance. A conducting element  512  is formed within the apparatus  300  to connect conductive elements  506  with the underside of dielectric  510 . The first pair of conductive elements  502  extend generally orthogonally to one another and are in electrical communication with a first pair of coaxial transmission lines  508  ( FIGS. 10 and 11 ) to form a high band waveguide having a first resonant frequency. A second pair of coaxial transmission lines  513  ( FIG. 11 ) are formed within the apparatus  300  electrically connected to the first pair of coaxial transmission lines  508  allowing the signal to be accessible on the ground side of waveguide transition module  500  at an isolated conductive pad. The apparatus  300  operates as a reflection surface for the first pair of conductive elements  502 .  FIG. 13  illustrates the return loss of the conductive elements  506  maximized at the resonant frequency. The insertion loss of the conductive elements  506  is minimized and balanced at the resonant frequency. The isolation of the conductive elements  506  is acceptable at the resonant frequency while maximized above the resonant frequency. 
     The second pair of conductive elements  506  are formed on a surface of a dielectric  510  and are similarly arranged orthogonal to one another and over a pair of conductive line sections  512  electrically connected to an isolated conductive pad on the underside of dielectric surface  510 . The conductive line sections  512  are generally orthogonal to the conductive elements  506 . The apparatus  300  operates as a reflection surface for the second pair of conductive elements  506 .  FIG. 13  illustrates the insertion loss, isolation and return loss of the dual band waveguide module  500 , with the losses Y 1  in dBs plotted relative to frequency in GHz. 
     Referring to  FIG. 14 , the various embodiments described herein may all be constructed as a multilayer wiring board using a plurality of high frequency laminate layers of material assembled to form a unitary printed wiring board.  FIG. 14  illustrates one exemplary printed wiring board  600  that includes a polar geometry bandgap portion  602 , which may be identical in construction to any of the above-described embodiments of  FIGS. 1-5 ,  7 ,  8 , or  10 - 12 . The printed wiring board  600  in this example includes a plurality of integrally formed layers such as a ground layer  604 , a layer  606  for clock signals, data layers  608  and  610 , a layer  612  on which a voltage is supplied, a “high band” layer  614  for stripline transmission of the high frequency signal, a voltage layer  616 , and a “low band” layer  618  for stripline transmission of the low frequency signal, and a ground layer  620 . In this example it will be noted that the entire printed wiring board  600  forms a stackup or single assembly having a height typically between 0.090 inch and 0.160 inch (2.286 mm and 4.06 m), which is significantly more compact than the 0.30 inch to 0.450 inch (7.62 mm-11.43 mm) height that is typically produced when using a quarter length back short assembly. The printed wiring board  600  also does not require the epoxying of two distinct circuit board sections together, as is required with a conventional quarter wavelength back short board. 
     The various embodiments thus provide high impedance bandgap surfaces and waveguide transition modules that can be manufactured significantly more easily and with less cost that traditional assemblies employing a quarter wavelength backshort. The various embodiments described herein can be used to produce an integrally formed printed wiring board assembly that is also significantly more compact that previous designs that require a quarter wavelength backshort. The various embodiments described herein are expected to find particular utility in a wide variety of phased array antenna applications for both civilian and military applications. 
     While various embodiments have been described, those skilled in the art will recognize modifications or variations which might be made without departing from the present disclosure. The examples illustrate the various embodiments and are not intended to limit the present disclosure. Therefore, the description and claims should be interpreted liberally with only such limitation as is necessary in view of the pertinent prior art.