Abstract:
An output circuit for a class AB push-pull amplifier includes an upper cascode output stage and a lower cascode output stage. The upper cascode stage includes first and second PMOS transistors connected in series between a positive power supply node and an output node, the first PMOS transistor configured to receive a first complementary input signal. The lower cascode output stage includes first and second NMOS transistors connected in series between a negative power supply node and the output node, the first NMOS transistor configured to receive a second complementary input signal. The output circuit also includes a bias circuit configured for providing a first bias voltage to a gate node of the second NMOS transistor and a second bias voltage to a gate node of the second PMOS transistor, in which the first and the second bias voltages being substantially proportional to the output voltage.

Description:
CROSS-REFERENCES TO RELATED APPLICATIONS 
     Not Applicable 
     BACKGROUND OF THE INVENTION 
     The present invention relates generally to CMOS integrated circuit techniques. More specifically, embodiments of the present invention provide methods and circuits for protecting amplifier output circuits. 
     Amplifier circuits are ubiquitous in modern electronic devices. An electronic amplifier increases the power and/or amplitude of a signal. In many applications, power amplifier circuits are used at the output stage of a system to drive an external device. Merely as an example, in an audio system, an output power amplifier is often used to drive an external speaker or headphone. 
     Power amplifier circuits output stages can be classified as class A, B, AB, and C, etc. for analog signal amplification. This classification is based on the portion of the input signal cycle during which the amplifying device conducts. 
     A Class A amplifier operates over the whole of the input cycle such that the output signal is a magnified replica of the whole input with no clipping. Class A amplifiers are the usual means of implementing small-signal amplifiers. In a Class A circuit, the amplifying device operated over the linear portion of its characteristic curve. Because the device is always conducting, even if there is no input at all, power is drawn from the power supply. Accordingly, class A amplifiers tend to be relatively inefficient, especially for large power devices. 
     In contrast, Class B amplifiers only amplify half of the input signal cycle. As such they tend to create signal distortion, but their efficiency is greatly improved over Class A amplifiers. This is because the amplifying element is switched off and does not dissipate power half of the time. An application using Class B amplifiers is the complementary pair or “push-pull” arrangement. Here, complementary devices are used to each amplify the opposite halves of the input signal. The amplified two halves are then recombined at the output. This arrangement gives improved efficiency, but can suffer from the drawback of mismatch at the “joints” between the two halves of the signal, also known as the crossover distortion. An improvement can be achieved by biasing the devices such that neither of the two devices is completely off when they&#39;re not in use. This mode of circuit operation is often called Class AB operation. 
     In Class AB operation, each device operates over half the wave similar to Class B operation, but each also conducts over a small signal range in the other half. As a result, when the waveforms from the two devices are combined, the crossover distortion is reduced. Here the two active elements conduct more than half of the time as a means to reduce the cross-over distortions of Class B amplifiers. 
     In certain applications, it may be desirable to use Class C amplifiers, which conduct less than 50% of the input signal and the distortion at the output is high, but high efficiencies are possible. An application for Class C amplifiers is in RF transmitters. 
     An audio amplifier is an electronic amplifier that amplifies low-power audio signals to drive loudspeakers. Audio signals generally have frequencies between 20 Hertz to 20,000 Hertz, which is the human range of hearing. In a typical audio system, the audio amplifier is usually preceded by low power audio amplifiers which perform tasks like pre-amplification, equalization, tone control, mixing/effects, or audio sources like record players, CD players, and mp3 streams. Audio systems are used in public address systems, theatrical and concert sound reinforcement, and home sound systems, and mobile phones and tablets etc. The sound card in a personal computer often contains several audio amplifiers, as does every stereo or home-theatre system. Audio amplifiers often need to meet stringent performance requirement. In some applications, the input signal to an audio amplifier may measure only a few hundred microwatts. However, its output power may be tens or hundreds of watts. 
     Because of these requirements, Class AB push-pull circuits are a popular design choice in audio power amplifiers. Even though audio amplifier circuits are widely used in many applications, certain limitations still exist. Some examples are discussed below.  FIG. 1A  is a simplified view diagram illustrating an output portion  100  of a conventional audio system. As shown in  FIG. 1A , an audio frequency signal  102  entering an amplifier  104 , which amplifies the signal and drives a speaker  108 . A schematic diagram of  100  is shown in  FIG. 1B , where the amplifier is shown as a preamplifier  105  and a CMOS output driver circuit  106  including a PMOS driver device and an NMOS driver device. The speaker  108  is shown as an equivalent 8 ohm resistance load. 
     In some class AB amplifiers, a cascode output stage is used. A cascode amplifier usually has a common source amplifier as input stage driven by signal source. This input stage drives a common gate amplifier as output stage. The cascode configuration would offer a potentially greater gain and much greater bandwidth. It also enables the use of low voltage devices in the higher voltage circuit. This is the main reason to use a cascode in an output stage. 
       FIG. 2  is a circuit diagram of a conventional output circuit for a push-pull class AB cascode amplifier. As shown in  FIG. 2 , output circuit  200  includes first power node  201  for coupling to a positive power supply V 0 , second power node  202  for coupling to a negative power supply V 1 , and an output node  205 . Output circuit  200  also includes first PMOS transistor P 1  and second PMOS transistor P 2  connected in series between positive power supply node  201  and output node  205 . The drain node of P 1  and the source node of P 2  are connected at node  207 . Output circuit  200  further includes first and second NMOS transistors N 1  and N 2  connected in series between output node  205  and negative power supply node  202 . The drain node of N 1  and the source node of N 2  are connected at node  208 . A first input node In 1  is coupled to a gate of the first PMOS transistor P 1 . A second input node In 2  is coupled to a gate of the first NMOS transistor N 1 . In  FIG. 2 , positive power supply V 0  and negative power supply V 1  are connected to a ground terminal GND. It can be seen in  FIG. 2  that the gate of PMOS transistor P 2  and the gate of NMOS transistor N 2  are both biased at a ground voltage GND. 
     BRIEF SUMMARY OF THE INVENTION 
     The inventor has observed that conventional cascode amplifiers suffer from various limitations. For example, conventional cascode devices are often biased with a constant voltage, halfway the supply voltage (or ground). These cascode devices may be adequate for quiescent operation and small output signals. However, this configuration can only handle large signals if the devices have a higher breakdown voltage than half the maximum voltage swing. This is undesirable because devices with higher breakdown voltages often require more complicated processes and higher cost. Therefore, cascode amplifier designs that use devices having low breakdown voltages, but allow large operating voltage range are highly desirable. 
     Embodiments of the invention provide a method to protect the output devices of a class AB output stage by providing bias voltages for the cascode transistors that depend on the output voltage. According to embodiments of the present invention, an output circuit for an amplifier includes a first power node for coupling to a positive power supply, a second power node for coupling to a negative power supply, and an output node. The output circuit also includes first and second PMOS transistors connected in series between the first power node and the output node, and first and second NMOS transistors connected in series between the output node and the second power node. The output circuit also includes a first input terminal coupled to a gate of the first PMOS transistor and a second input terminal coupled to a gate of the first NMOS transistor. Moreover, the output circuit includes a voltage divider coupled between the output node and a ground node GND, the voltage divider including first and second resistors which are connected at a first node. The output circuit also include two source followers. A first source follower includes a third PMOS transistor, which has a gate coupled to the first node and a source coupled to a gate of the second NMOS transistor. A second source follower includes a third NMOS transistor, which has a gate coupled to the first node and a source coupled to a gate of the second PMOS transistor. 
     In an embodiment of the above output circuit, the first PMOS transistor is configured to receive a first input signal, and the first NMOS transistor is configured to receive a second input signal. 
     In an embodiment, a bias voltage at the gate of the second NMOS transistor is configured to follow a voltage at the output node. 
     In an embodiment, a drain voltage of the first NMOS transistor is determined by the resistance values of the first and the second resistors and the threshold voltages of the third PMOS transistor and the second NMOS transistor. 
     In an embodiment, the first NMOS transistor is characterized by a drain-to-source voltage that is below a first voltage limit during operation. 
     In an embodiment, a bias voltage at the gate of the second PMOS transistor is configured to follow a voltage at the output node. 
     In an embodiment, a drain voltage of the first PMOS transistor is determined by the resistance values of the first and the second resistors and the threshold voltages of the third NMOS transistor and the second PMOS transistor. 
     In an embodiment, the first PMOS transistor is characterized by a drain-to-source voltage that is below a second voltage limit during operation. 
     In an embodiment, a bias voltage at the gate of the second NMOS transistor is configured to follow a voltage at the output node. 
     According to alternative embodiments of the present invention, an output circuit for a class AB push-pull amplifier includes an upper cascode output stage including first and second PMOS transistors connected in series between a positive power supply node and an output node, in which the first PMOS transistor is configured to receive a first complementary input signal. The output circuit also has a lower cascode output stage including first and second NMOS transistors connected in series between a negative power supply node and the output node, in which the first NMOS transistor is configured to receive a second complementary input signal. The output circuit also has a bias circuit, which includes a voltage divider and two source followers. The voltage divider is coupled to the output node configured for providing a first voltage signal that is related to the voltage at the output node. A first source follower is coupled to receive the first voltage signal and is configured for providing a first bias voltage to a gate node of the second NMOS transistor in the lower cascode output stage. A second source follower is coupled to receive the first voltage signal and is configured for providing a second bias voltage to a gate node of the second PMOS transistor in the upper cascode output stage. 
     In an embodiment of the above output circuit, the first source follower includes a third PMOS transistor, the third PMOS transistor having a gate coupled to a first node of the voltage divider and a source coupled to a gate of the second NMOS transistor. 
     In an embodiment, the second source follower includes a third NMOS transistor, the third NMOS transistor having a gate coupled to a first node of the voltage divider and a source coupled to a gate of the second PMOS transistor. 
     In an embodiment, the first NMOS transistor is characterized by a drain-to-source voltage that is below a first voltage limit during operation. 
     In an embodiment, the first PMOS transistor is characterized by a drain-to-source voltage that is below a second voltage limit during operation. 
     According to alternative embodiments of the present invention, an output circuit for a class AB push-pull amplifier includes an upper cascode output stage and a lower cascode output stage. The upper cascode output stage includes first and second PMOS transistors connected in series between a positive power supply node and an output node, the first PMOS transistor configured to receive a first complementary input signal. The lower cascode output stage includes first and second NMOS transistors connected in series between a negative power supply node and the output node, the first NMOS transistor configured to receive a second complementary input signal. The output circuit also includes a bias circuit configured for providing a first bias voltage to a gate node of the second NMOS transistor and a second bias voltage to a gate node of the second PMOS transistor, in which the first and the second bias voltages being substantially proportional to the output voltage. 
     In an embodiment of the above output circuit, the bias circuit includes a voltage divider coupled to the output node configured for providing a first voltage signal that is proportional to the voltage at the output node. The bias circuit also includes a first source follower coupled to receive the first voltage signal and configured for providing the first bias voltage to the gate node of the second NMOS transistor. The bias circuit also has a second source follower coupled to receive the first voltage signal and configured for providing the second bias voltage to the gate node of the second PMOS transistor. 
     In an embodiment, the first source follower includes a third PMOS transistor, which has a gate coupled to a first node of the voltage divider and a source coupled to a gate of the second NMOS transistor. 
     In an embodiment, the second source follower includes a third NMOS transistor, the third NMOS transistor having a gate coupled to a first node of the voltage divider and a source coupled to a gate of the second PMOS transistor. 
     In an embodiment, the first NMOS transistor is characterized by a drain-to-source voltage that is below a first voltage limit during operation. 
     In an embodiment, the first PMOS transistor is characterized by a drain-to-source voltage that is below a second voltage limit during operation. 
     A further understanding of the nature and advantages of the present invention may be realized by reference to the remaining portions of the specification and the drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1A  is a schematic diagram of an output portion of a conventional audio system; 
         FIG. 1B  is a schematic diagram of an output portion of another conventional audio system; 
         FIG. 2  is a circuit diagram of a conventional output circuit for a cascoded push-pull class AB amplifier; 
         FIG. 3  is a diagram illustrating voltages at various circuit nodes in the output circuit in  FIG. 2 ; 
         FIG. 4  is another diagram illustrating voltages at various circuit nodes in the output circuit in  FIG. 2 ; 
         FIG. 5  is a circuit diagram of an output circuit according to an embodiment of the present invention; 
         FIG. 6  is a diagram illustrating voltages at various circuit nodes in the output circuit in  FIG. 5  according to an embodiment of the present invention; and 
         FIG. 7  is another diagram plotting voltages at various circuit nodes of  FIG. 5  as the output voltage varies according to an embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     In an audio system with a charge pump to generate a negative supply voltage, the amplifiers to drive the output such as the line out, the headphone, or an earpiece, voltages across output transistors can become higher than their breakdown voltages. Stacking (also called cascoding) devices helps to make the circuits more robust. As illustrated in  FIG. 2  and described above, conventional cascode devices are often biased with a constant voltage, halfway the supply voltage (or ground). This design may be adequate for quiescent operation and small output signals. However, as described below, with large signals, the voltage across the devices can be higher than the breakdown voltage limit of the transistors, thus requiring more costly devices with higher breakdown voltages. 
       FIG. 3  is a diagram illustrating the voltages at various circuit nodes in the output circuit in  FIG. 2 . In  FIG. 3 , the vertical axis shows the magnitude of the voltages at various circuit nodes, and the horizontal axis shows the input voltage in the output circuit in  FIG. 2 . In this example, the positive supply voltage can be as high as 2V, and the negative supply voltage as low as −2V.  FIG. 3  shows the voltages in the circuit when the input of the amplifier is changed from −2 to 2V, with amplifier gain is set to 1. In this example, the transistors have a source-drain voltage limit of 2V. Curve  301  shows the voltage at the output node  205  in  FIG. 2 , which, in principle, can vary between −2V and 2V. Curve  307  represents the voltage at the drain node  208  of NMOS transistor N 1 , which is also the voltage at the source node of NMOS transistor N 2 . It can be seen that the source-drain voltage drop of NMOS transistor N 2  can be as high as 2.25V, exceeding the desired specification of the source-drain voltage of 2V. 
       FIG. 4  is another diagram plotting the voltages at various circuit nodes of  FIG. 2  as the output voltage varies.  FIG. 4  is similar to  FIG. 3  described above, but with the focus on the PMOS cascode transistors P 1  and P 2  in  FIG. 2 . In  FIG. 4 , the vertical axis shows the magnitude of the voltages at various circuit nodes, and the horizontal axis shows the input voltage in the output circuit in  FIG. 2  with the gain of the class AB amplifier set to 1. Curve  301  shows the voltage at the output node  205 , which, in principle, can vary between −2V and 2V. Curve  407  represents the voltage at the drain node  207  of PMOS transistor P 1 , which is also the voltage at the source node of PMOS transistor P 2 . It can be seen that the source-drain voltage drop of PMOS transistor P 2  can be as high as 2.47V, exceeding the desired specification of the source-drain voltage of 2V. 
     It can be seen that in the conventional output circuit, the source-to-drain voltages of the cascode transistors often cannot be maintained within the device specification of the transistors. Embodiments of the invention provide a method to protect the output devices of a class AB output stage by providing bias voltages for the cascode transistors that depend on the output voltage such that the voltage across the cascode transistors remain within the voltage specification during operation. In an embodiment, the bias circuit includes a voltage divider connected to the output of the amplifier and two source followers that are connected between the voltage divider and the two cascode transistors. An example is described below. 
       FIG. 5  is a circuit diagram of an output circuit according to an embodiment of the present invention. As shown in  FIG. 5 , output circuit  500  includes first power node  501  for coupling to a positive power supply V 0 , second power node  502  for coupling to a negative power supply V 1 , and an output node  505 . Output circuit  500  also includes first PMOS transistor P 1  and second PMOS transistor P 2  connected in series between positive power supply node  501  and output node  505 . The drain node of P 1  and the source node of P 2  are connected at node  507 . Output circuit  500  further includes first and second NMOS transistors N 1  and N 2  connected in series between output node  505  and negative power supply node  502 . The drain node of N 1  and the source node of N 2  are connected at node  508 . A first input node In 1  is coupled to a gate of the first PMOS transistor P 1 . A second input node In 2  is coupled to a gate of the first NMOS transistor N 1 . A voltage divider  510  is coupled to output node  505  and includes first resistor R 1  and second resistor R 2 . The first and the second resistors R 1  and R 2  are connected at a first internal divider node  515 . Output circuit  500  also includes a first source follower  520 , which includes a third PMOS transistor P 3  having a gate coupled to the first internal divider node  515  and a source node  522  coupled to a gate of the second NMOS transistor N 2  for providing a bias. Output circuit  500  also includes a second source follower  530 , which includes a third NMOS transistor N 3  having a gate coupled to the first internal divider node  515  and a source node  532  coupled to a gate of the second PMOS transistor P 2  for providing a bias. 
     As shown in  FIG. 5 , the substrate nodes of PMOS transistors P 1  and P 2  are coupled to positive power node  501 . The substrate nodes of NMOS transistors N 1  and N 2  are coupled to negative power node  502 . Source follower PMOS transistor P 3  has a substrate node coupled to its source node. Similarly, source follower NMOS transistor N 3  has a substrate node coupled to its source node. In  FIG. 5 , input node In 1  is coupled to output node  505  through serially connected resistor R 3  and capacitor C 1 . Similarly, input node In 2  is coupled to output node  505  through serially connected resistor R 4  and capacitor C 2 . 
     In  FIG. 5 , positive power supply V 0  is connected between positive power node  501  and a ground node GND, and negative power supply V 1  is connected between ground terminal GND and negative power node  502 . Source follower  520  also includes a load device PMOS transistor P 4  coupled to PMOS transistor P 3 . PMOS transistor P 4  has a gate connected to GND, and a source and a substrate connected to positive power node  501 . Source follower  530  also includes a load device NMOS transistor N 4  coupled to NMOS transistor N 3 . NMOS transistor P 4  has a gate connected to GND, and a source and a substrate connected to negative power node  502 . 
     In an embodiment, output circuit  500  in  FIG. 5  can be used as an output device for a class AB amplifier. Input nodes In 1  and In 2  are configured to receive input signals, e.g., audio input signals, and provide the input signals to cascode output transistors P 1 , P 2 , N 1 , and N 2 . As shown in  FIG. 5 , the bias voltages of transistors P 2  and N 2  are derived from a bias circuit and are dependent on the output voltage at output node  505 . The bias circuit includes voltage divider  510  connected to the output of the amplifier, and two source followers  520  and  530  that are connected between the voltage divider and the two cascode transistors P 2  and N 2 . Here, the voltage divider allows for a fraction on of the output voltage to be used as the bias voltage. In addition, the source followers can provide isolation in the bias circuit. 
     In some embodiments, the drain nodes of transistors P 3  and N 3  are coupled to GND to provide higher gate-source voltage Vgs at cascode transistors N 2  and P 2 . In other embodiments, the drain nodes of transistors P 3  and N 3  are not necessarily coupled to GND. 
       FIG. 6  is a diagram illustrating the voltages at various circuit nodes in the output circuit in  FIG. 5  according to an embodiment of the present invention. In  FIG. 6 , the vertical axis shows the magnitude of the voltages at various circuit nodes, and the horizontal axis shows the input voltage in the output circuit in  FIG. 5 , with the gain of the class AB amplifier set at 1. In this example, the positive supply voltage can be as high as 2V, and the negative supply voltage as low as −2V. In this embodiment, the transistors have a source-drain voltage limit of 2V. Curve  601  shows the voltage at the output node  505 , which can vary between −2V and 2V. In this embodiment, however, a clamping circuit, not shown in  FIG. 5 , prevents the output voltage from reaching the limits of 2V and −2V at the far ends of the graph. In  FIG. 5 , curve  603  shows the voltage at the first internal divider node  515  of the voltage divider which, through voltage divider resistors R 1  and R 2 , follows curve  601 , the voltage at the output node  505 . Curve  605  shows the gate voltage of NMOS transistor N 2  in  FIG. 5 . It can be seen that the gate bias voltage  605  at cascode transistor N 2  follows the voltage at the first internal divider node  515  of the voltage divider, which in turn follows curve  601 , the voltage at the output node. Curve  607  represents the voltage at the drain node  508  of NMOS transistor N 1 , which is also the voltage at the source node of NMOS transistor N 2 . It can be seen that the drain node  508  of NMOS transistor N 1  is kept below 0V. As a result, drain voltage of NMOS transistor N 1  is kept between −2V and 0V. Accordingly, the source-drain voltage drop of NMOS transistor N 1  is kept no more than 2V, meeting the desired specification of the source-drain voltage of 2V. Moreover, the source-drain voltage drop of NMOS transistor N 2 , which is the voltage between curves  601  and  607 , is also kept within 2V. 
     With reference to the circuit diagram of  FIG. 5  and the voltage diagrams in  FIG. 6 , it can be seen that the drain voltage  607  of NMOS transistor N 1  is the same as the source voltage of NMOS transistor N 2 , which is below the gate voltage of transistor N 2  by the gate-source voltage of transistor N 2 . From  FIG. 5 , it can be seen that the gate bias of transistor N 2  is derived from the first internal divider node  515  of voltage divider  410  by a difference of the gate-source voltage of source follower PMOS transistor P 3 . The gate-source voltage of a transistor in turn is related to its threshold voltage, and the voltage at the first internal divider node  515  is derived from the output voltage at output node  505  through voltage divider  510 . Therefore, in embodiments of the present invention, the drain voltage of cascode NMOS transistor N 2  can be determined by the voltage divider resistors R 1  and R 2 , and the threshold voltages of NMOS cascode transistor N 2  and PMOS source follower transistor P 3 . By selecting appropriate values for the resistances of resistors R 1  and R 2 , and the threshold voltages for transistors N 2  and P 3 , the voltage drop across the drain-source nodes of transistors N 1  and N 2  can be maintained within a desired voltage specification. Further, using a source follower in the bias circuit can serve to isolate the gate bias of the cascode transistor from the output node. 
     It is also noted that in  FIG. 6  that curve  605 , representing the gate voltage of NMOS transistor N 2 , doesn&#39;t go below the ground voltage. With a resistive load, NMOS transistors N 1  and N 2  will have to sink a large current when the output voltage is close to the negative supply rail. The on resistance of N 2  should be as small as possible. Therefore, the gate voltage of N 2  is clamped to ground, and does not follow the output voltage  601  to the negative voltage range. 
       FIG. 7  is another diagram illustrating the voltages at various circuit nodes of  FIG. 5  as the output voltage varies according to an embodiment of the present invention.  FIG. 7  is similar to  FIG. 6  described above, but with the focus on the PMOS cascode transistors P 1  and P 2 . In  FIG. 7 , the vertical axis shows the magnitude of the various voltages, and the horizontal axis shows the input voltage in the output circuit in  FIG. 5 , with the gain of the class AB amplifier set at 1. Again, the positive supply voltage is 2V, and the negative supply voltage is −2V. The transistors have a source-drain voltage limit of 2V. Similar to those shown in  FIG. 6 , curve  601  shows the voltage at the output node  505 , and curve  603  shows the voltage at the internal node  515  of the voltage divider, which follows curve  601 , the voltage at the output node. In  FIG. 7 , curve  706  shows the gate bias voltage of PMOS transistor P 2  in  FIG. 5 . It can be seen that the gate bias voltage at cascode transistor P 2  follows the voltage at the internal node  515  of the voltage divider, which in turn follows curve  601 , the voltage at the output node  505 . In  FIG. 7 , curve  708  represents the voltage at the drain node of PMOS transistor P 1 , which is also the voltage at the source node of PMOS transistor P 2 . It can be seen that the drain node  507  of PMOS transistor P 1  is kept above 0V. As a result, drain voltage of PMOS transistor P 1  is kept between −2V and 0V. Accordingly, the source-drain voltage drop of PMOS transistor P 1  is kept no more than 2V, meeting the desired specification of the source-drain voltage of 2V. Moreover, the source-drain voltage drop of PMOS transistor P 2 , which is the voltage between curves  601  and  708 , is also kept within 2V. 
     It is also noted that in  FIG. 7  that curve  706 , representing the gate voltage of PMOS transistor P 2 , does not go above the ground voltage. With a resistive load, PMOS transistors P 1  and P 2  will have to source a large current when the output voltage is close to the positive supply rail. The on resistance of P 2  should be as small as possible. Therefore, the gate voltage of P 2  is clamped to ground, and does not follow the output voltage  601  to the high voltage range. 
     Similar to the description above in connection with  FIG. 5 , by selecting appropriate values for the resistances of resistors R 1  and R 2 , and the threshold voltages for cascode transistors P 2  and source follower transistor N 3 , the voltage drop across the drain-source nodes of transistor P 1  can be maintained within a desired voltage specification. 
     Thus, embodiments of the invention provide a method for protecting the output devices of a class AB output stage by providing bias voltages for the cascode transistors that depend on the output voltage. In an embodiment, the bias circuit includes a voltage divider connected to the output of the amplifier and two source followers that are connected between the voltage divider and the two cascode transistors. 
     Various embodiments of the present invention are described above. It is understood that the examples and embodiments described herein are for illustrative purposes only and that various modifications or changes in light thereof will be suggested to persons skilled in the art and are to be included within the spirit and purview of this application and scope of the appended claims.