Abstract:
A fiber-optic measurement device ( 10 ) includes a SAGNAC ring interferometer ( 20 ) having a proper frequency f p . The aim is to improve response time while maintaining high precision across the measurement range. Biasing elements ( 130 ) are used to produce: a first biasing phase-differential modulation component Δφ b1 (t) ( 34 ) which is periodic in time-slots, having levels +π and −π, at a first biasing modulation frequency f b1  such that f b1 =(2k 1 +1) f p , k 1  being a natural number; and a second periodic biasing phase-differential modulation component Δφ b2 (t) ( 35 ), having extreme amplitudes +π/a and −π/a, a being a non-zero real number such that |a|&gt;1, at a second biasing modulation frequency f b2  such that f b2 =(2k 2 +1) f p , k 2  being a non-zero natural number such that k 2 &gt;k 1 . A gyrometer including such a measurement device and a navigation or inertial-stabilization system including at least one such gyrometer are also described.

Description:
FIELD OF THE INVENTION 
     The invention relates to a fiber-optic measurement device allowing to measure the variation of a parameter that produces non-reciprocal disturbances in a SAGNAC ring interferometer. 
     BACKGROUND OF THE INVENTION 
     The SAGNAC interferometer and the physical phenomena involved thereby are well known. Reference may be made for example about that to “The Fiber-Optic Gyroscope”, H. Lefèvre (Artech House, 1993). 
     In such an interferometer, a splitting plate or any other splitting device splits an incident wave at the input of the interferometer into two waves. The two thus-created waves are referred to as “counter-propagating waves”. They indeed propagate in opposite directions along a same closed optical path, then recombine with each other, producing interferences at the time of their recombination. The interference state between the two counter-propagating waves then depends on the relative phase difference between them. The luminous power P measured at the output of a SAGNAC interferometer is of the form: P(Δφ)=P 0 [1+cos(Δφ)], where Δφ is the relative phase difference between the two counter-propagating waves. Hence, the power measured at the output of the interferometer takes values between a minimum (it is then talked about “dark” fringe) and a maximum (“bright” fringe) as a function of the value of the phase difference Δφ. 
     It is known that some physical phenomena are liable to introduce so-called non-reciprocal phase shifts, in the counter-propagating waves, hence generating a phase difference Δφ p  between these waves and modifying the interference state during the recombination thereof. Hence, the measurement of this non-reciprocal phase shift Δφ p  allows to quantify the phenomenon that has been generated thereby. 
     The main physical phenomenon liable to create non-reciprocal disturbances is the SAGNAC effect produced by the rotation of the interferometer about an axis perpendicular to the plane of its closed optical path. A second effect, the FARADAY effect—or collinear magneto-optic effect—is also known for producing non-reciprocal effects of this type. 
     It is known that a SAGNAC interferometer can include a fiber-optic coil, which is preferably single-mode and of the polarization-maintaining type. The multiple turns of an optical fiber form a closed optical path of very long length, up to several kilometers. 
     A proper frequency f p  of the SAGNAC interferometer is commonly defined. The proper frequency f p  of a SAGNAC ring interferometer including a single-mode fiber-optic coil (silica fiber having a refractive index close to 1.5 in the operating wavelength range) of 1 kilometer long is of the order of 100 kilohertz (kHz). The extension of the coil length and hence of the optical path has for advantage to provide the interferometer with a greater sensitivity. 
     Moreover, it has been shown that the measurement accuracy is improved by the use of a so-called “phase cancellation” method, also called closed-loop operation, instead of a simple open-loop operation. 
     According to this method, an additional so-called “feedback” phase difference Δφ cr  is introduced by means of a phase modulator between the two counter-propagating waves, so as to compensate for the phase shift Δφ p  produced by the parameter measured. The sum of the two phase-shifts Δφ p  and Δφ cr  is kept at zero, which allows to make the interferometer operate with a better accuracy. The measurement of the parameter to be measured is made thanks to the use of the signal necessary to the production of the feedback phase difference Δφ cr . 
     However, the sensitivity of the response P(Δφ) of the interferometer in the vicinity of the zero phase difference (Δφ=0) is low, because the signal measured at the output of the interferometer is a cosine-wave function of the phase difference Δφ. 
     It is known that it is possible to displace the operating point of the interferometer towards a point offering a greater sensitivity. It has notably been proposed to introduce an additional so-called “biasing” phase-difference modulation Δφ b , by means of the phase modulator. The total phase difference Δφ t  between the two counter-propagating waves is then equal to the sum of the different phase differences: Δφ t =Δφ p +Δφ cr +Δφ b . 
     A simple-to-implement solution to perform this biasing consists in a square-wave periodic modulation at a biasing modulation frequency f b , the modulation having for example levels +π/2 and −π/2. This biasing phase-difference modulation Δφ b  produces at the output of the interferometer a square-wave periodic modulated electrical signal at the biasing modulation frequency f b  whose amplitude is a sine-wave function of the sum of the two phase-shifts Δφ p  and Δφ cr , in the case of a closed-loop measurement as described above. The response provided by the SAGNAC interferometer can hence be used with a greater sensitivity. 
     Moreover, in order to improve the stability of the measurement of a non-reciprocal parameter by means of a SAGNAC interferometer, the document EP0430747 proposes a device in which the biasing phase-difference modulation MOO introduced between the two counter-propagating waves is periodic at the frequency f b . 
     At each period of the modulation, the level of the phase-difference modulation Δφ b (t) is hence equal to:
         φ 0  during the first quarter of period,   aφ 0  during the second quarter of period,   −φ 0  during the third quarter of period, and   −aφ 0  during the fourth quarter of period.       

     The values of a and φ 0  are chosen so as to verify the relation: cos(φ 0 )=cos(aφ 0 ). 
     The device according to the document EP0430747 also includes a signal processing system using the four values taken by the modulated electrical signal delivered by the interferometer during one modulation period. The signal processing system then allows to maintain constant the gain of the modulation chain so as to compensate for the slow drifts of the different components of the device (for example: variation as a function of the temperature). 
     To reduce the effects of the modulation chain defects on the measurement, it is known that the biasing modulation frequency f b  has to be equal to the proper frequency f p  of the interferometer or to one of its odd multiples. 
     In particular, the so-called “four states” modulation generated by the biasing described in the document EP0430747 introduces defect-bearing peaks on the modulated electrical signal measured at the output of the interferometer, these defects being eliminated when the biasing modulation frequency f b  is equal to the proper frequency f p  of the SAGNAC interferometer, or to one of its odd multiples. 
     Moreover, the number of these peaks increases with the biasing modulation frequency f b . 
     To reduce the response time of the interferometer and to ensure that the feedback loop of measurement does not break in case of a rapid variation of the parameter to be measured, the biasing modulation frequency f b  is increased. However, the measurement accuracy is hence substantially degraded due to a greater number of peaks in the signal detected. 
     SUMMARY OF THE INVENTION 
     The object of the present invention is to propose a fiber-optic measurement device wherein a parameter to be measured generates a phase difference between two counter-propagating waves, wherein the response time is improved while keeping a good accuracy over the measurement range. 
     For that purpose, the invention relates to a fiber-optic measurement device of the type in which a parameter to be measured generates a phase difference Δφ p  between two counter-propagating waves, including:
         a light source,   a fiber-optic SAGNAC ring interferometer, preferably single-mode, including a coil and a splitting element, in which said two counter-propagating waves propagate, said ring interferometer having a proper frequency f p ,   an electromagnetic radiation detector, receiving the luminous power exiting from said ring interferometer and delivering a modulated electrical signal representative of the luminous power, which is function of the total phase difference Δφ t  between said two counter-propagating waves at the output of said ring interferometer,   a modulation chain adapted to modulate said luminous power exiting from said ring interferometer, said modulation chain including:
           at least one digital/analog converter adapted to process a digital control signal to deliver an analog control signal,   an amplifier adapted to process said analog control signal to deliver a modulation control voltage V m (t),   at least one phase modulator placed in said interferometer, which, when subjected at the input to said modulation control voltage V m (t), is adapted to generate at the output a phase-shift modulation φ m (t), said phase-shift modulation φ m (t) introducing between said two counter-propagating waves a phase-difference modulation Δφ m (t) such that: Δφ m (t)=φ m (t)−φ m (t−Δτ g ), Δτ g =1/(2f p ) being the transit-time difference between said two counter-propagating waves determined between said phase modulator and said splitting element, and   
           signal processing means including:
           an analog/digital converter digitizing said modulated electrical signal received from the detector and representative of said power received to deliver a digital electrical signal, and   a digital processing unit adapted to process said digital electrical signal to deliver a signal function of said phase difference Δφ p  and of said parameter to be measured,   
           feedback means adapted to process said signal function of said phase difference Δφ p  to generate a feedback signal,   biasing means adapted to generate a biasing signal,   means for controlling said modulation chain, adapted to process said feedback signal and said biasing signal to deliver said digital control signal at the input of said modulation chain, such that said modulation control voltage V m (t) at the input of said phase modulator is the sum of a feedback modulation voltage V cr (t) and a biasing modulation voltage V b (t), said phase modulator being adapted, when it is subjected to said feedback modulation voltage V cr (t), to generate a feedback phase-shift modulation φ cr (t), said feedback phase-shift modulation φ cr (t) introducing a feedback phase-difference modulation Δφ cr  between said two counter-propagating waves allowing to keep at zero the sum of said phase difference Δφ p  and said feedback phase-difference modulation Δφ cr , and   means for controlling the gain of said modulation chain allowing to keep adjusted the transfer function of said modulation chain.       

     According to the invention, said fiber-optic measurement device is characterized in that said biasing means are adapted to generate said biasing signal such that said phase modulator generates a biasing phase-shift modulation φ b (t), when it is subjected to said biasing modulation voltage V b (t), said biasing phase-shift modulation φ b (t) being the sum of:
         a first biasing phase-shift modulation component φ b1 (t), introducing a first biasing phase-difference modulation component Δφ b1 (t) between said two counter-propagating waves, said first biasing phase-difference modulation component Δφ b1 (t) being a square-wave periodic modulation, of levels +π and −π, at a first biasing modulation frequency f b1  such that f b1 =(2k 1 +1)f p , k 1  being a natural number and f p  said proper frequency, and   a second biasing phase-shift modulation component φ b2 (t), introducing a second biasing phase-difference modulation component Δφ b2 (t) between said two counter-propagating waves, said second biasing phase-difference modulation component Δφ b2 (t) being a periodic modulation, of extreme amplitudes +π/a and −π/a, a being a non-zero real number such that |a|&gt;1, at a second biasing modulation frequency f b2  such that f b2 =(2k 2 +1)f p , k 2  being a non-zero natural number such that k 2 &gt;k 1 , and f p  said proper frequency.       

     Hence, said device according to the invention allows to perform a “biasing” around π, thanks to said first biasing phase-difference modulation component Δφ b1 (t) offering an optimal signal-to-noise ratio for the detection chain. Said device also allows to increase the frequency of demodulation of the signal function of said phase difference Δφ p  and of said parameter to be measured thanks to said second biasing phase-difference modulation component Δφ b2 (t) to reduce the response time of the interferometer without thereby increasing the number of cumbersome peaks in the modulated electrical signal. The stability of the closed loop is hence improved, that is to say that the measurement device according to the invention is capable of measuring a parameter generating a non-reciprocal effect, even if said parameter to be measured varies very rapidly. 
     Moreover, other advantageous and non-limitative characteristics of the device according to the invention are as follows:
         said first biasing phase-difference modulation component Δφ b1 (t) has a duty factor of 50%;   said second biasing phase-difference modulation component Δφ b2 (t) has a duty factor of 50%;   the first biasing modulation frequency f b1  of said first biasing phase-difference modulation component Δφ b1 (t) is such that k 1 =0;   the second biasing modulation frequency f b2  of said second biasing phase-difference modulation component Δφ b2 (t) is such that k 2 &gt;2;   the second biasing modulation frequency f b2  of said second biasing phase-difference modulation component Δφ b2 (t) is such that k 2 &gt;4;   the second biasing modulation frequency f b2  of said second biasing phase-difference modulation component Δφ b2 (t) is a square-wave modulation;   the second biasing modulation frequency f b2  of said second biasing phase-difference modulation component Δφ b2 (t) is a sine-wave modulation;   the second biasing modulation frequency f b2 =(2k 2 +1)f p  is such that f b2 =(2k 21 +1)f b1 , k 21  being a non-zero natural number, and f b1 =(2k 1 +1)f p  being the first biasing modulation frequency, and said first biasing phase-difference modulation component Δφ b1 (t) and said second biasing phase-difference modulation component Δφ b2 (t) are in phase quadrature;   said feedback phase-shift modulation φ cr (t) is a stair-step modulation;   said feedback phase-shift modulation φ cr (t) and said first biasing phase-shift modulation component φ b1 (t) are synchronous, each stair-step of said feedback phase-shift modulation φ cr (t) having a duration Δτ g  and said first biasing phase-shift modulation component φ b1 (t) being a first biasing modulation frequency f b1  such that f b1 =f p , f p  being the proper frequency;   said feedback phase-shift modulation φ cr (t) falls down by 2π when it exceeds 2π;   said digital processing unit demodulates said digital electrical signal in phase with said second biasing phase-difference modulation component Δφ b2 (t) independently of the first biasing phase-difference modulation component Δφ b1 (t), and said means for controlling the gain of said modulation chain demodulate said digital electrical signal so as to provide a signal function of the transfer function of said modulation chain.       

     The measurement device according to the invention is particularly well adapted for the making of a gyrometer. In this case, the parameter to be measured is a component of the rotational speed of the ring interferometer. 
     Hence, the invention also relates to a gyrometer, characterized in that it is compliant with the fiber-optical measurement device according to the invention, the parameter to be measured being a component of the rotational speed of the ring interferometer. 
     This gyrometer advantageously enters into the making of navigation or inertial-stabilization systems. 
     Hence, the invention also proposes a navigation or inertial-stabilization system including at least one gyrometer according to the invention. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Embodiments of the invention will be described in detail with reference to the drawings in which: 
         FIG. 1  shows a schematic view of the measurement device according to the prior art; 
         FIG. 2  shows a functional scheme representing the different means implemented in the measurement device according to the prior art; 
         FIG. 3  shows an example according to the prior art of a stair-step feedback phase-shift modulation φ cr (t) falling down by 2π when it exceeds 2π; 
         FIG. 4  shows an example according to a particular embodiment of the invention of first and second biasing phase-difference modulation components Δφ b1 (t) and Δφ b2 (t); 
         FIG. 5  shows an example according to a particular embodiment of the invention of first and second biasing phase-shift modulation components φ b1 (t) and φ b2 (t), generating the first and second biasing phase-difference modulation components Δφ b1 (t) and φ b2 (t) shown in  FIG. 4 ; 
         FIG. 6  shows an example of phase-difference modulation Δφ m (t) according to a particular embodiment of the invention, the open-loop total phase difference in the interferometer and the corresponding modulated electrical signal produced by the detector when the parameter to be measured generates a zero phase difference Δφ p  and when the transfer function of the modulation chain is correctly adjusted; 
         FIG. 7  shows an example of phase-difference modulation Δφ m (t) according to a particular embodiment of the invention, the open-loop total phase difference in the interferometer and the corresponding modulated electrical signal produced by the detector when the parameter to be measured generates a non-zero phase difference Δφ p  and when the transfer function of the modulation chain is correctly adjusted; 
         FIG. 8  shows an example of phase-difference modulation Δφ m (t) according to a particular embodiment of the invention, the open-loop total phase difference in the interferometer and the corresponding modulated electrical signal produced by the detector when the parameter to be measured generates a zero phase difference Δφ p  and when the transfer function of the modulation chain is incorrectly adjusted; 
         FIG. 9  shows an example of phase-difference modulation Δφ m (t) according to a particular embodiment of the invention, the open-loop total phase difference in the interferometer and the corresponding modulated electrical signal produced by the detector when the parameter to be measured generates a non-zero phase difference Δφ p  and when the transfer function of the modulation chain is incorrectly adjusted. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       FIG. 1  shows a fiber-optic measurement device  10  according to the prior art, of type in which a parameter to be measured generates a phase difference Δφ p  between two waves. 
     The fiber-optic measurement device  10  first includes a light source  11  herein comprising a laser diode. 
     As a variant, the light source may comprise, for example, a super-luminescent diode or a doped-fiber light source of the “ASE” (“Amplified Spontaneous Emission”) type. 
     The device  10  also comprises a first splitting element  12 . This first splitting element  12  is herein a semi-reflective plate having a transmittance of 50% and a reflectance of 50%. 
     As a variant, the splitting element may, for example, be a 3-decibel 2×2 coupler or an optical circulator. 
     The luminous wave emitted by the light source  11  is hence transmitted in part by the first splitting element  12  towards an optical filter  13  at the output of which the luminous wave has been filtered. The optical filter  13  preferably includes a polarizer and a spatial filter. This spatial filter is herein a single-mode optical fiber, preferably of the polarization-maintaining type. 
     The device  10  also includes a SAGNAC ring interferometer  20  comprising a fiber-optic coil  21  wound around itself. It is herein an optical fiber, preferably of the single-mode and polarization-maintaining type. 
     This SAGNAC ring interferometer  20  also comprises a second splitting element  23  allowing to split the wave exiting from the optical filter  13  into two counter-propagating waves  24 ,  25  on the two “arms” of the ring interferometer  20 , these two arms defining two optical paths  24 A and  25 A. The second splitting element  23  is herein a semi-reflective plate having a transmittance of 50% and a reflectance of 50%. 
     The second splitting element  23  also allows to recombine the two counter-propagating waves  24 ,  25  at the output of the ring interferometer  20 . 
     As a variant, the second splitting element may be, for example, a 3-decibel 2×2 coupler or a “Y”-junction in integrated optics. 
     The two counter-propagating waves  24 ,  25  then pass through the optical filter  13  and are reflected by the first splitting element  12  towards an electromagnetic radiation detector  14 . 
     This detector  14  is herein a semi-conductor photodiode. 
     The detector  14  is sensitive to the luminous power received, which is herein function of the interference state between the two counter-propagating waves  24 ,  25  during their recombination at the output of the SAGNAC ring interferometer  20 . It hence delivers an electrical signal that is representative of the total phase difference Δφ t  between the two counter-propagating waves  24 ,  25 . It will be seen in the following of the description that this electrical signal is a modulated electrical signal. 
     The device  10  also includes a modulation chain  30  comprising a digital/analog converter  31 , an amplifier  32  and a phase modulator  33 . 
     The digital/analog converter  31  processes a digital control signal delivered by the control means  140 , the decomposition of this signal being described in detail hereinafter. The digital/analog converter  31  delivers as an output an analog control signal. 
     The amplifier  32  then processes this analog control signal to deliver a modulation control voltage V m (t) to the phase modulator  33 . 
     The phase modulator  33  is placed in the ring interferometer  20  and is thus also a part thereof. It is herein arranged at one end of the optical path of the SAGNAC ring interferometer  20 . The phase modulator  33  is herein of the electro-optical type (said of “Pockels effect” type) in proton-exchange lithium-niobate integrated optics. 
     The phase modulator  33  allows, when the time-dependant control modulation voltage V m (t) is applied at the input thereof, to generate a phase-shift modulation φ m (t) that is proportional, and thus with the same time dependency, on a luminous wave passing through it at the given instant t in one direction or another. 
     In the case of the SAGNAC ring interferometer  20  shown in  FIG. 1 , the transit-time difference of the counter-propagating waves  24 ,  25  along the two optical paths  24 A,  25 A between the phase modulator  33  and the second splitting element  23  is denoted Δτ g . 
     Hence, the phase-shift modulation φ m (t) generated by the phase modulator  33  controlled by the modulation control voltage V m (t) introduces between the two counter-propagating waves  24 ,  25 , a phase-difference modulation Δφ m (t) such that: Δφ m (t)=φ m (t)−φ m (t−Δτ g ). 
     The transit-time difference Δτ g  also defines a proper frequency f p  of the SAGNAC ring interferometer  20  by the relation: f p =1/(2Δτ g ). 
     This proper frequency f p  thus depends on the length of the coil  21  in the SAGNAC ring interferometer  20 . With the fiber-optic coil  21  used herein, a coil having a length of 1 kilometer, the proper frequency f p  of the SAGNAC ring interferometer  20  is of about 100 kilohertz (kHz), corresponding to a transit-time difference Δτ g  of 5 microseconds (μs). 
     The luminous power P(Δφ 1 ) received by the detector  14  is also modulated and the electrical signal delivered by the detector  14  will thus be a modulated electrical signal ( 38 ), examples of which are given in  FIGS. 6 to 9 . 
     The modulated electrical signal  38  is transmitted to electronic means  100  that process it to deliver a signal function of the phase difference Δφ p  and of the parameter to be measured. 
     For that purpose, the electronic means  100  comprise signal processing means  110 , as shown in  FIG. 2 . These signal processing means  110  include an analog/digital converter  111  digitizing the modulated electrical signal  38  provided by the detector  14  to deliver a digital electrical signal. 
     This digitization operation is performed at a synchronization frequency fixed by the clock  101 . The synchronization frequency of the clock  101  is preferably a multiple of the proper frequency f p  of the SAGNAC ring interferometer  20 . 
     The signal processing means  110  also comprise a digital processing unit  112  configured to process the digital electrical signal provided at the output of the analog/digital converter  111 . The digital processing unit  112  also includes a digital demodulator, a control-loop digital filter fed with a first demodulated digital signal exiting from the digital demodulator and a register. 
     The digital processing unit  112  delivers a signal function of the phase difference Δφ p  and of the parameter to be measured for any desired external use. 
     The electronic means  100  also control in return the modulation chain  30 . 
     For that purpose, the electronic means  100  include, on the one hand, feedback means  120  and, on the other hand, biasing means  130 . 
     The feedback means  120  receive as an input the signal function of the phase difference Δφ p  of the parameter to be measured provided by the digital processing unit  112 . The feedback means  120  generate as an output a feedback signal whose action on the modulation chain  30  will be described in more detail hereinafter. 
     The feedback means  120  herein include an accumulator. 
     The biasing means  130  are configured to generate a biasing signal at precise instants, synchronized by the frequency of the clock  101 . The action of this biasing signal on the modulation chain  30  will be described in more detail hereafter. 
     The electronic means  100  further include control means  140  that have two inputs and one output. The control means  140  receive as an input, on the one hand, the feedback signal, and on the other hand, the biasing signal. These signals are then processed by the control means  140 . The control means  140  deliver as an output a digital control signal that is then transmitted to the digital/analog converter  31  of the modulation chain  30 . 
     The control means  140  herein include a digital adder. The operation performed by the control means  140  consists in the digital addition of the feedback signal provided by the feedback means  120  and of the biasing signal provided by the biasing means  130 . The digital control signal is the signal resulting from this addition. 
     The digital control signal is then transmitted to the modulation chain  30 . It is converted into an analog control signal by the digital/analog converter  31 , then transmitted to the amplifier  32  that delivers a modulation control voltage V m (t) to the phase modulator  33 . 
     The modulation chain  30  thus receives as a input the digital control signal and produces as an output, by means of the phase modulator  33 , a phase-shift modulation φ m (t) modulated in time, which will be introduced in the counter-propagating waves  24 ,  25  propagating in the SAGNAC ring interferometer  20 . 
     The modulation chain  30  is then characterized electronically by its transfer function between the input and the output. This transfer function is the ratio between the value (in radians) of the phase-shift effectively generated by the modulation chain  30  via the phase modulator  33  and the value (with no unity) of the digital control signal transmitted to the modulation chain  30 . 
     In order to keep the transfer function of the modulation chain  30  adjusted, the electronic means  100  also include gain control means  150 . 
     These gain control means  150  include another digital processing unit (not shown) using the digital electrical signal delivered by the analog/digital converter  111  so as to provide a signal function of the transfer function of the modulation chain  30 . 
     This signal is filtered by a control-loop digital integrating filter that feeds another digital/analog converter controlling the variable gain G of the amplifier  32  or the analog reference voltage of the digital/analog converter  31 . Hence, the transfer function of the modulation chain  30  is kept correctly adjusted, as well as the modulation control voltage V m (t) delivered by the amplifier  32  to the phase modulator  33 . 
     It is meant by this that a given value of the digital control signal at the input of the modulation chain  30  will always give the same value (in radians) of phase-shift modulation φ m  generated by the phase modulator  33 , and hence the same value (in radians) of the phase-difference modulation Δφ m  introduced between the two counter-propagating waves  24 ,  25  in the SAGNAC ring interferometer  20 . 
     The digital control signal being the sum of the feedback signal and the biasing signal, the modulation control voltage V m (t) at the input of the phase modulator  33  is decomposed into the sum of a feedback modulation voltage V cr (t) and a biasing modulation voltage V b (t). 
     The feedback modulation voltage V cr (t) at the input of the phase modulator  33  results, at the output of the phase modulator  33 , in a feedback phase-shift modulation φ cr (t) on the wave passing through it. 
     In the case of the SAGNAC ring interferometer  20 , the effect of this feedback phase-shift modulation φ cr (t) is the introduction of a feedback phase-difference modulation Δφ cr (t) between the two counter-propagating waves  24 ,  25 , allowing to compensate for the phase difference Δφ p  generated by the parameter to be measured and hence to keep at zero the sum of the phase difference Δφ p  and the phase difference Δφ cr . 
     This feedback allows to make the device  10  operate in closed loop so as to reach a good linearity and stability of the measurement of the parameter generating the phase difference Δφ p . 
     According to the prior art,  FIG. 3  relates to the feedback phase-shift modulation φ cr (t) generated by the phase modulator  33  from the feedback modulation voltage V cr (t). 
     The feedback signal generated by the feedback means  120  is a stair-step digital signal. 
     For a SAGNAC ring interferometer  20 , of proper frequency f p , the prior art teaches a duration of Δτ g  for each step, the passage from one step to another being synchronized thanks to the clock  101  present in the electronic means  100 . 
     This is translated in  FIG. 3  at the level of the feedback phase-shift modulation φ cr (t) that is a stair-step modulation. As mentioned hereinabove, the prior art teaches that the duration of the steps of the feedback phase-shift modulation φ cr (t) is equal to Δτ g . 
     Likewise, as described hereinabove, the height of the step is such that the phase-difference modulation Δφ cr (t) introduced between the two counter-propagating waves  24 ,  25  compensate for the phase difference Δφ p  due to the parameter measured. 
     Moreover, the feedback phase-shift modulation φ cr (t) is a stair-step ramp modulation such that this modulation falls down by 2π, as shown in  FIG. 3 , when the value of the step exceeds 2π. 
     It is known that this “falling down to 2π” is made necessary by the fact that the value of the feedback modulation voltage V cr (t) cannot increase indefinitely. The use of digital means, such as the digital/analog converter  31 , allows to make simply this falling down to 2π. 
     As mentioned above, the biasing means  130  are configured to generate a biasing signal, this biasing signal being transmitted to the control means  140  piloting the modulation chain  30 . 
     This biasing signal is associated with the biasing modulation voltage V b (t), through the digital/analog converter  31  and the amplifier  32 . 
     This biasing modulation voltage V b (t) at the output of the amplifier  32  and at the input of the phase modulator  33  results at the output of the phase modulator  33  in a biasing phase-shift modulation φ b (t) on a wave passing through it. 
     In the case of the SAGNAC ring interferometer  20 , the effect of this biasing phase-shift modulation φ b (t) is the introduction of a biasing phase-difference modulation Δφ b (t) between the two counter-propagating waves  24 ,  25 . 
     The modulation control voltage V m (t) at the input of the phase modulator  33  being decomposed into the sum of the feedback modulation voltage V cr (t) and the biasing modulation voltage V b (t), the phase-shift modulation φ m (t) (respectively the phase-difference modulation Δφ m (t)) is the sum of the feedback phase-shift modulation φ cr (t) (respectively the feedback phase-difference modulation Δφ cr (t)) and the biasing phase-shift modulation φ b (t) (respectively the biasing phase-difference modulation Δφ b (t)), such that: φ m (t)=φ cr (t)+φ b (t), and Δφ m (t)=Δφ cr (t)+Δφ b (t). 
     According to the invention, the biasing phase-shift modulation φ b (t) is the sum of:
         a first biasing phase-shift modulation component φ b1 (t), and   a second biasing phase-shift modulation component φ b2 (t).       

     For that purpose, the biasing means  130  are arranged so that the biasing digital signal is the sum of a first biasing component and a second biasing component. 
     The first biasing component is associated with a first voltage component V b1 (t), through the digital/analog converter  31  and the amplifier  32 . 
     Likewise, the second biasing component is associated with a second voltage component V b2 (t), through the digital/analog converter  31  and the amplifier  32 . 
     Therefore, the biasing modulation voltage V b (t) is decomposed into the sum of a first voltage component V b1 (t) and a second voltage component V b2 (t), generated through the amplifier  32  and the digital/analog converter  31 , respectively from the first biasing component and the second biasing component. 
     According to the invention, the first biasing phase-shift modulation component φ b1 (t)  35 A, generated from the first voltage component V b1 (t) thanks to the phase modulator  33 , introduces a first biasing phase-difference modulation component Δφ b1 (t) between the counter-propagating waves  24 ,  25  of the SAGNAC ring interferometer  20 . 
     According to the invention, this first biasing phase-difference modulation component Δφ b1 (t) is a square-wave periodic modulation at a first biasing modulation frequency f b1  such that f b1 =(2k 1 +1)f p , k 1  being a natural number and f p  the proper frequency. 
     According to a preferred embodiment of the invention, the first biasing modulation frequency f b1  is herein equal to the proper frequency f p (k 1 =0) of the SAGNAC ring interferometer  20 . 
     As a variant, the first biasing modulation frequency f b1  may be, for example, an odd multiple of the proper frequency f p , such that k 1 &gt;0. 
     An example of this first biasing phase-difference modulation component Δφ b1 (t) is shown and denoted by  34  in  FIG. 4 . 
     According to the invention, the first biasing phase-difference modulation component Δφ b1 (t)  34  has extreme levels of values +π and −π. This first modulation component will hence be referred to hereinafter “π-modulation”. 
     In a preferred embodiment of the invention, this π-modulation, denoted by  34 , has herein a duty factor of 50%, i.e. the duration of the level +π (respectively the level −π) represents 50% (respectively 50%) of the total duration of the period of π-modulation, denoted by  34 . 
     The first biasing modulation frequency f b1  being herein equal to the proper frequency f p =1/(2Δτ g ) of the SAGNAC ring interferometer  20 , the period of the π-modulation, denoted by  34 , is equal to 1/f b1 =2Δτ g , the π-modulation, denoted by  34 , remaining at its level +π during a half-period of duration Δτ g , and at its level −π during another half-period of duration Δτ g . 
     The first voltage component V b1 (t) produces a first biasing phase-shift modulation component φ b1 (t)  34 A as shown in  FIG. 5 . The phase modulator  33  is a reciprocal modulator, the SAGNAC ring interferometer  20  behaves as a delay line between the two counter-propagating waves  24 ,  25 , such that the first biasing phase-difference modulation component Δφ b1 (t)  34  verifies the relation: Δφ b1 (t)=φ b1 (t)−φ b1 (t−Δτ g ). 
     Therefore, it is understood how the first biasing phase-difference modulation component Δφ b1 (t)  34 , shown in  FIG. 4 , is generated from the first biasing phase-shift modulation component φ b1 (t)  34 A, shown in  FIG. 5 . 
     In a particular embodiment of the invention, where each stair step of the feedback phase-shift modulation φ cr (t) has a duration Δτ g , the feedback phase-shift modulation φ cr (t) created through the modulation chain  30  is synchronous with the first biasing phase-shift modulation component φ b1 (t) which is herein at the proper frequency f p . 
     It will be defined herein that the feedback phase-shift modulation φ cr (t) and the first biasing phase-shift modulation component φ b1 (t) are in phase with each other when the passage of the feedback phase-shift modulation φ cr (t) from one step to another occurs during a transition of the first biasing phase-shift modulation component φ b1 (t) from one extreme level to another. 
     According to this particular embodiment, the falling down to 2π of the feedback phase-shift modulation φ cr (t) is then synchronized with a transition of the π-modulation, denoted by  34 . 
     According to the invention, the second biasing phase-shift modulation component φ b2 (t), generated from the second voltage component V b2 (t) thanks to the phase modulator  33 , introduces a second biasing phase-difference modulation component φ b2 (t) between the counter-propagating waves  24 ,  25  of the SAGNAC ring interferometer  20 . 
     According to the invention, this second biasing phase-difference modulation component φ b2 (t) is a periodic modulation at a second biasing modulation frequency f b2 , such that f b2 =(2k 2 +1)f p , k 2  being a non-zero natural number such that k 2 &gt;k 1  and f p  being the proper frequency. 
     The second biasing modulation frequency f b2  is hence a frequency strictly higher than the first biasing modulation frequency f b1 . 
     According to a particular embodiment of the invention, the second biasing modulation frequency f b2  is such that f b2 =3f p  (i.e. k 2 =1). It is hence effectively strictly higher than the first biasing modulation frequency f b1 , which is herein such that f b1 =f p . 
     In another embodiment, the second biasing modulation frequency f b2  is preferentially such that k 2 &gt;2, and still more preferentially such that k 2 &gt;4. 
     According to a particular embodiment of the invention, the second biasing phase-difference modulation component Δφ b2 (t) is herein a square-wave modulation. 
     An example of this second biasing phase-difference modulation component Δφ b2 (t) is shown and denoted by  35  in  FIG. 4 . It can be observed that the second biasing phase-difference modulation component Δφ b2 (t)  35  has, in this example, extreme levels of values +π/8 and −π/8. The second modulation component of this example will hence be referred to hereinafter “π/8-modulation”. 
     Generally, the second biasing phase-difference modulation component Δφ b2 (t) may have extreme levels of values +π/a and −π/a, a being a real number verifying the condition |a|&gt;1. The second modulation component is then generally referred to as “π/8-modulation”. 
     As shown in  FIG. 4 , the π/8-modulation, denoted by  35 , has preferably a duty factor of 50%, i.e. the duration of the level +π/8 (respectively the level −π/8) represents 50% (respectively 50%) of the total duration of the period of modulation π/8, denoted by  35 . 
     The second biasing modulation frequency f b2  being herein equal to 3f p =3/(2Δτ g ) of the SAGNAC ring interferometer  20 , the period of π/8-modulation, denoted by  35 , is equal to 1/f b2 =(2/3)Δτ g , the π/8-modulation, denoted by  35 , remaining at its level +π/8 during a half-period of duration (1/3)Δτ g , and at its level −π/8 during another half-period of duration (1/3)Δτ g . 
     According to another embodiment of the invention, the second biasing phase-difference modulation component Δφ b2 (t) is a sine-wave periodic modulation, of amplitude π/a, such that a is a non-zero real number verifying the condition |a|&gt;1. 
     The second biasing modulation voltage V b2 (t) produces a second biasing phase-shift modulation component φ b2 (t)  35 A as shown in  FIG. 5 . By analogy with the first modulation component described hereinabove, the second biasing phase-difference modulation component Δφ b2 (t)  35  verifies the relation: Δφ b2 (t)=φ b2 (t)−φ b2 (t−Δτ g ). 
     Therefore, it is understood how is generated the second biasing phase-difference modulation component Δφ b2 (t)  35 , shown in  FIG. 4 , from the second biasing phase-shift modulation component φ b2 (t)  35 A, shown in  FIG. 5 . 
     According to the particular embodiment described hereinabove, the second biasing modulation frequency f b2  is an odd multiple of the first biasing modulation frequency f b1 . Indeed, the first biasing modulation frequency f b1  being such that f b1 =f p , the second biasing modulation frequency f b2  is such that f b2 =3f p =3f b1 =(2k 21 +1)f b1 , with k 21 =1. 
     Moreover, the first biasing phase-difference modulation component Δφ b1 (t)  34  and the second biasing phase-difference modulation component Δφ b2 (t)  35  are herein in phase quadrature. 
     It will be defined herein that the first biasing phase-difference modulation component Δφ b1 (t)  34  and the second biasing phase-difference modulation component Δφ b2 (t)  35  are in phase quadrature when a transition of the first biasing phase-difference modulation component Δφ b1 (t)  34  from one extreme level to another one occurs at equal distance from two successive zeroes of the second biasing phase-difference modulation component Δφ b2 (t)  35 . 
     As illustrated in  FIG. 4 , the modulation π, denoted by  34 , operates a transition from the level +π to the level −π at the instant t=t 1 . Likewise, the π/8-modulation, denoted by  35 , is cancelled at two instants t 2  and t′ 2  about the considered transition of the modulation π, denoted by  34 . The modulation π, denoted by  34 , and the π/8-modulation, denoted by  35 , are in phase quadrature because herein |t 1 −t 2 |=t 1 −t′ 2 |. 
     In  FIGS. 6 to 9 , different operations of a particular embodiment of the device according to the invention are shown:
         in  FIG. 6 , the phase difference Δφ p  generated by the parameter to be measured is zero and the transfer function of the modulation chain  30  is correctly adjusted,   in  FIG. 7 , the phase difference Δφ p  generated by the parameter to be measured is non-zero and the transfer function of the modulation chain  30  is correctly adjusted,   in  FIG. 8 , the phase difference Δφ p  generated by the parameter to be measured is zero and the transfer function of the modulation chain  30  is incorrectly adjusted, and   in  FIG. 9 , the phase difference Δφ p  generated by the parameter to be measured is non-zero and the transfer function of the modulation chain  30  is incorrectly adjusted.       

     In each of the  FIGS. 6 to 9 , where Δφ=Δφ b +Δφ p  represents the open-loop phase difference at the output of the ring interferometer  20  and t represents the time, it has been shown:
         the biasing phase-difference modulation Δφ b (t)  36 , which is the sum of the first biasing phase-difference modulation component Δφ b1 (t)  34  and the second biasing phase-difference modulation component Δφ b2 (t)  35 ,   the luminous power P(Δφ)  37  received by the detector  14 , and   the modulated electrical signal S(t)  38  delivered by the detector  14 .       

     For more simplicity, the reasoning is made for these  FIGS. 6 to 9  on a device  10  placed in open loop. In this case, the luminous power P(Δφ)  37  received by the detector  14  is a cosine-wave function of the relative phase difference Δφ between the two counter-propagating waves  24 ,  25  in the SAGNAC ring interferometer  20 . 
     The luminous power P(Δφ)  37  received by the detector  14  is indeed of the form: P(Δφ)=P 0 [1+cos(Δφ)]. It is hence zero when Δφ=+π or −π, (because cos(+π)=cos(−π)=−1) and it is maximum and equal to 2P 0  when Δφ=0 (because cos(0)=1). 
     The reasoning may be transposed to the case of the closed loop. 
     The first biasing phase-difference modulation component Δφ b1 (t)  34  (modulation π) being herein at the frequency f p  and the second biasing phase-difference modulation component Δφ b2 (t)  35  (π/8-modulation) being at the frequency 3f p , the biasing phase-difference modulation Δφ b (t)  36 , which is the sum of the two previous modulations, is hence a periodic modulation at the proper frequency f p . 
     As described hereinabove, the modulation π, denoted by  34 , has two extreme levels +π and −π and the π/8-modulation, denoted by  35 , has two extreme levels +π/8 and −π/8, so that the biasing phase-difference modulation Δφ b (t) has sequentially four different levels defining four different modulation states, which are:
         State E1 or state “&lt;+−&gt;”: when the modulation π, denoted by  34 , is at its “high” extreme level +π and when the π/8-modulation, denoted by  35 , is at its “low” extreme level −π/8,   State E2 or state “&lt;++&gt;”: when the modulation π, denoted by  34 , is at its “high” extreme level +π and when the π/8-modulation, denoted by  35 , is at its “high” extreme level +π/8,   State E3 or state “&lt;−+&gt;”: when the modulation π, denoted by  34 , is at its “low” extreme level −π and when the π/8-modulation, denoted by  35 , is at its “high” extreme level +π/8,   State E4 or state “&lt;−−&gt;”: when the modulation π, denoted by  34 , is at its “low” extreme level −π and when the π/8-modulation, denoted by  35 , is at its “low” extreme level −π/8,       

     These four distinct modulation states E1, E2, E3, E4 are preferably close to a dark fringe of the SAGNAC ring interferometer  20 , where the signal-to-noise ratio is optimum. 
     The luminous power P(Δφ)  37  received by the detector  14  is hence modulated according to these four distinct modulation states and the modulated electrical signal S(t)  38  delivered by the detector  14  takes sequentially four values S1, S2, S3 and S4 respectively associated with the four modulation states E1, E2, E3 and E4. 
     When the parameter to be measured generates a zero phase difference Δφ p , as it is the case in  FIG. 6 , then the four levels of the biasing phase-difference modulation Δφ b (t)  36  are:
         For the state E1 (state &lt;+−&gt;):
 
Δφ=Δφ b +Δφ p =Δφ b =Δφ b1 +Δφ b2 =+π−π/8=8π/8
   For the state E2 (state &lt;++&gt;):
 
Δφ=Δφ b +Δφ p =Δφ b =Δφ b1 +Δφ b2 =+π+π/8=9π/8
   For the state E3 (state &lt;−+&gt;):
 
Δφ=Δφ b +Δφ p =Δφ b =Δφ b1 +Δφ b2 =−π+π/8=−7π/8
   For the state E4 (state &lt;−−&gt;):
 
Δφ=Δφ b +Δφ p =Δφ b =Δφ b1 +Δφ b2 =−π−π/8=−9π/8
       

     The luminous power P(Δφ)  37  received by the detector  14  being a cosine-wave function, as explained above, it is herein, in the case of  FIG. 6 , the same whatever the modulation state. The detector  14  hence delivers a modulated electrical signal S(t)  38  taking four identical values S1, S2, S3 and S4. 
     From the preceding situation, described in  FIG. 6 , where the parameter to be measured generates a zero phase-difference Δφ p , the situation passes to that described in  FIG. 7 , where the parameter to be measured generates a non-zero phase difference Δφ p  between the two counter-propagating waves  24 ,  25  in the SAGNAC ring interferometer  20 . It will be considered, in the example of  FIG. 7 , that the phase difference Δφ p  generated by the parameter to be measured is of π/16. 
     This may be shown in  FIG. 7  by “offsetting” the biasing phase-difference modulation Δφ b (t)  36  by the value Δφ p . This offset generates a change of the four modulation states on which is modulated the signal received by the detector  14 , which is function of the luminous power P(Δφ)  37  received, depending on the total phase difference Δφ at the output of the ring interferometer  20 . The four levels of the biasing phase-difference modulation Δφ b (t)  36  associated with the four modulation states are hence now:
         For the state E1 (state &lt;+−&gt;):
 
Δφ=Δφ b +Δφ p =[+π−π/8]+π/16=15π/16
   For the state E2 (state &lt;++&gt;):
 
Δφ=Δφ b +Δφ p =[+π+π/8]+π/16=19π/16
   For the state E3 (state &lt;−+&gt;):
 
Δφ=Δφ b +Δφ p =[−π+π/8]+π/16=−13π/16
   For the state E4 (state &lt;−−&gt;):
 
Δφ=Δφ b +Δφ p =[−π−π/8]+π/16=−17π/16
       

     Hence, as can be seen in  FIG. 7 , the luminous power P(Δφ)  37  received by the detector  14  in the modulation states E1 and E4 is lower and that received in the modulation state E2 and E3 is higher. 
     The detector  14  then delivers a modulated electrical signal S(t)  38  as shown in  FIG. 7 . This modulated electrical signal S(t)  38  takes sequentially the four values S1, S2, S3 and S4 respectively associated with the four states of modulation E1, E2, E3 and E4. These four values S1, S2, S3 and S4 taken by the modulated electrical signal S(t)  38  are herein identical two-by-two: S1=S4 and S2=S3. 
     It is also observed in  FIG. 7  that the modulated electrical signal S(t)  38  has peaks  39  corresponding alternately to the transitions from the state E1 to the state E4 of modulation and from the state E3 to the state E2 of modulation, when the luminous power P(Δφ) received passes by a maximum at the value Δφ=0. 
     These peaks  39  are cumbersome insofar as they introduce non wanted defects in the modulated electrical signal S(t)  38 . 
     This modulated electrical signal S(t)  38  is then digitized by the analog/digital converter  111  that delivers and transmits a digital electrical signal to the digital processing unit  112 . 
     This digital electrical signal is also modulated and takes four digital values Σ1, Σ2, Σ3 and Σ4 according to the four modulation states E1, E2, E3 and E4 of the biasing phase-difference modulation Δφ b (t)  36  with which the values are associated. 
     The digital processing unit  112  demodulates the digital electrical signal in phase with the second biasing phase-difference modulation component Δφ b2 (t)  35  independently of the first biasing phase-difference modulation component Δφ b1 (t)  34 . 
     It is meant by this that the digital processing unit  112  delivers a first demodulated digital signal Σ p  based on the four digital values Σ1, Σ2, Σ3 and Σ4 respectively associated with the four modulation states E1, E2, E3 and E4, by performing a calculation operation of the type: Σ p =−Σ1+Σ2+Σ3−Σ4 where the “weight” of each digital value in the preceding expression depends on the extreme level of the π/8-modulation, denoted by  35 , in the modulation state associated with this digital value, but does not depend on the level of the modulation π, denoted by  34 , in this modulation state. 
     The digital processing unit  112  hence produces a first demodulated digital signal Σ p  depending of the phase-shift Δφ p  and representative of the value of the parameter to be measured in the SAGNAC ring interferometer  20 . 
     It is herein observed that the first demodulated digital signal Σ p  is a signal at the same frequency that the π/8-modulation, denoted by  35 , i.e. 3f p . Nevertheless, it is also observed that the defect-bearing peaks  39  in the modulated electrical signal S(t)  38  transmitted by the detector  14  occur at the frequency equal to the double of the frequency of the modulation π, denoted by  34 , i.e. herein at the frequency 2f p . 
     Hence, it is herein possible, by using a first biasing phase-difference modulation component Δφ b1 (t)  34  at the frequency f p  and a second biasing phase-difference modulation component Δφ b2 (t)  35  at the frequency 3f p , to obtain a signal representative of the parameter to be measured at a frequency 3f p  and to limit the number of defect-bearing peaks  39  in the modulated electrical signal S(t)  38  delivered by the detector  14 . The device  10  may then detect the rapid variations of the parameter to be measured, without thereby degrading the accuracy of the measurement. 
     Generally, according to the invention, the number of defect-bearing peaks  39  is function of the first biasing modulation frequency f b1  of the first biasing phase-difference modulation component Δφ b1 (t)  34  and the frequency of the signal function of the parameter to be measured provided by the digital processing unit  112  is equal to the second biasing modulation frequency f b2  of the second biasing phase-difference modulation component Δφ b2 (t)  35 . Hence, the response time of the fiber-optic measurement device  10  according to the invention is substantially reduced and the accuracy of the measurement is maintained. 
     In a closed-loop operation, the first demodulated digital signal Σ p  serves as an error signal to control the total phase difference Δφ t  to zero by compensating the non-reciprocal phase-shift Δφ p  with the opposite phase-shift Δφ cr  introduced by the phase modulator  33  controlled by the feedback means  120 . 
     This phase-shift Δφ cr  being generated through the same modulation chain  30  as the biasing phase-difference modulation Δφ b , the control of the modulation chain  30 , whose operation is described in detail hereinafter, thus allows to have a stable and controlled measurement of Δφ cr , and hence finally of Δφ p , which is opposite thereto and which is the parameter that is desired to be measured. 
       FIG. 8  shows the case of a fiber-optic measurement device  10  according to a particular embodiment of the invention, wherein the measured parameter generates a zero phase difference Δφ p , and wherein the transfer function of the modulation chain  30  is incorrectly adjusted. As above, the reasoning is herein made in open loop. 
     In practice, the transfer function, which depends on the characteristics of both the digital/analog converter  31  via its analog reference voltage and the amplifier  32  via its variable gain G, may undergo variations as a function of the measurement conditions, for example the temperature of operation of the device  10  of the electrical drift of certain electronic components included in the electronic means  100 . Generally, the parameters influencing the transfer function generate low and slow variations of the latter, so that the gain control means  150  operate easily and rapidly so as to keep adjusted the transfer function of the modulation chain  30 . 
     In the case shown in  FIG. 8 , the first biasing phase-difference modulation component Δφ b1 (t)  34  is a square-wave periodic modulation, of extreme levels (1+ε)·[π]=16π/15 and (1+ε)·[−π]=−16π/15, the parameter ε being a quantity representative of the deviation with respect to the transfer function of the correctly-adjusted modulation chain  30 . Herein, the transfer function of the modulation chain  30  is such that the parameter ε is 1/15. 
     On the other hand, the first biasing modulation frequency f b1  of the first biasing phase-difference modulation component Δφ b1 (t)  34  remains unchanged and herein equal to the proper frequency f p . 
     Likewise, as shown in  FIG. 8 , the second biasing phase-difference modulation component Δφ b2 (t)  35  is herein a square-wave modulation, of extreme levels (1+ε)·[π/8]=2π/15 and (1+ε)·[−π/8]=−2π/15, that is periodic at a second biasing modulation frequency f b2  remaining unchanged and equal to 3f p . 
     These changes of extreme levels of modulation has for consequence that the biasing phase-difference modulation Δφ b (t)  36  is also modified by the multiplicative factor (1+ε)=16/15. 
     Hence, this homothetic transformation on the four modulation levels causes a change of the four modulation states E1, E2, E3 and E4 on which is modulated the signal received by the detector  14 , which is function of the luminous power P(Δφ)  37  received depending on the open-loop phase difference Δφ the output of the SAGNAC ring interferometer  20 . 
     The four levels of the biasing phase-difference modulation Δφ b (t)  36  associated to the four modulation states are hence now:
         For the state E1 (state &lt;+−&gt;):
 
Δφ=Δφ b +Δφ p =(1+ε)[π−π/8]=14π/15
   For the state E2 (state &lt;++&gt;):
 
Δφ=Δφ b +Δφ p =(1+ε)[π+π/8]=18π/15
   For the state E3 (state &lt;−+&gt;):
 
Δφ=Δφ b +Δφ p =(1+ε)[−π+π/8]=−14π/15
   For the state E4 (state &lt;−−&gt;):
 
Δφ=Δφ b +Δφ p =(1+ε)[−π−π/8]=−18π/15
       

     Hence, the luminous power P(Δφ)  37  received by the detector  14  in the modulation states E1 and E4 is identical, but lower than the luminous power received when the transfer function of the modulation chain  30  is correctly adjusted, as in  FIGS. 7 and 8 . 
     Likewise, the luminous power P(Δφ)  37  received by the detector  14  in the modulation states E2 and E3 is identical, but higher than the luminous power received when the transfer function of the modulation chain  30  is correctly adjusted, as in  FIGS. 7 and 8 . 
     The detector  14  then delivers a modulated electrical signal S(t)  38  as shown in  FIG. 8 . This modulated electrical signal S(t)  38  takes sequentially four values S1, S2, S3 and S4 respectively associated with the four modulation states E1, E2, E3 and E4. These four values are herein identical two-by-two: S1=S3 and S2=S4. 
     The four values Σ1, Σ2, Σ3 and Σ4 of the digital electrical signal respectively associated with the four modulation states E1, E2, E3 and E4 being also identical two-by-two with Σ1=Σ3 and Σ2=Σ4, the first demodulated digital signal Σ p , calculated according to the operation Σ p =−Σ1+Σ2+Σ3−Σ4, is zero, which indicates that the value of the phase difference Δφ p  due to the parameter to be measured is also zero. 
     Moreover, the digital electrical signal delivered by the analog/digital converter  111  is transmitted to the gain control means  150  as shown in  FIG. 2 . 
     The gain control means  150  demodulate the digital electrical signal so as to provide a signal function of the transfer function of the modulation chain  30 . 
     More precisely, the other digital processing unit of the gain control means  150  performs a calculation operation of the type: Σ G =Σ1−Σ2+Σ3−Σ4, so as to produce a second demodulated digital signal Σ G  independent of the phase difference Δφ p  generated by the parameter to be measured but significant of the transfer function of the modulation chain  30 . 
     In particular, in the case shown in  FIG. 8 , the second demodulated digital signal Σ G  is non-zero, the transfer function of the modulation chain  30  being incorrectly adjusted. 
     The second demodulated digital signal Σ G  then serves as an error signal for a control loop of the transfer function of the modulation chain  30 . 
     For that purpose, the second demodulated digital signal Σ G  is filtered by a control-loop digital integrating filter that then feeds the digital/analog converter  31  to control the analog reference voltage or the amplifier  32  to control the variable gain G thereof. 
     Hence, the transfer function of the modulation chain  30  is kept correctly adjusted between the value of the digital control signal and the value of the phase-shift modulation effectively applied by the phase modulator  33 . 
     It will be observed that, in the case of  FIGS. 6 and 7 , the second demodulated digital signal Σ G  is zero because the transfer function of the modulation chain  30  is correctly adjusted. 
     Indeed, in this case:
         Σ1=Σ4, the luminous power P(Δφ)  37  received in the state E1 and in the state E4 being the same, and   Σ2=Σ3, the luminous power P(Δφ)  37  received in the state E2 and in the state E3 being the same.       

       FIG. 9  shows the case of a fiber-optic measurement device  10  according to a particular embodiment of the invention, wherein the measured parameter generates a non-zero phase difference Δφ p , herein equal to π/24 and wherein the transfer function of the modulation chain  30  is incorrectly adjusted. As above, the reasoning is herein made in open loop. 
     In the particular case of  FIG. 9 , the transfer function of the modulation chain  30  is herein also such that ε=1/15. 
     Hence, the four levels of the biasing phase-difference modulation Δφ b (t)  36  associated with the four modulation states are hence herein:
         For the state E1 (state &lt;+−&gt;):
 
Δφ=Δφ b +Δφ p =(1+ε)[π−π/8]+π/24=117π/120
   For the state E2 (state &lt;++&gt;):
 
Δφ=Δφ b +Δφ p =(1+ε)[π+π/8]+π/24=149π/120
   For the state E3 (state &lt;−+&gt;):
 
Δφ=Δφ b +Δφ p =(1+ε)[−π+π/8]+π/24=−107π/120
   For the state E4 (state &lt;−−&gt;):
 
Δφ=Δφ b +Δφ p =(1+ε)[−π−π/8]+π/24=−139π/120
       

     As can be observed in  FIG. 9 , the modulated electrical signal S(t)  38  then takes four values S1, S2, S3 and S4, which are all different. 
     Hence, the first demodulated digital signal Σ p  representative of the phase difference Δφ p  and of the parameter to be measured in the SAGNAC ring interferometer  20  is non-zero. 
     Likewise, the second demodulated digital signal Σ G  significant of the transfer function of the modulation chain  30  is non-zero, showing that the latter is effectively incorrectly adjusted. 
     Hence, the measurement of the phase difference Δφ p  due to the parameter to be measured and that of the transfer function of the modulation chain ( 30 ) are performed independently of each other. 
     The measurement device of the invention is particularly well adapted to the making of a gyrometer. In this case, the parameter to be measured is a component of the rotational speed of the ring interferometer  20 . 
     This gyrometer hence advantageously enters in the making of navigation or inertial-stabilization systems. 
     Such an arrangement is also well adapted to the making of a magnetic-field and electrical-current measurement device, taking advantage of the FARADAY effect.