Abstract:
Several embodiments of a block downconverter using a SBAR bandpass filter in a superheterodyne receiver are disclosed. The block downconverter is coupled to receive a radio frequency input that includes a target region. The block downconverter is configured to produce a selected one of an overlapping plurality of portions of the target region as an intermediate frequency (IF) block having a fixed center frequency. Furthermore, the block downconverter includes a semiconductor bulk acoustic resonator (SBAR) filter that operates as an IF filter.

Description:
FIELD OF THE INVENTION  
         [0001]    This invention relates to signal analyzers, and more particularly to signal analyzers for radio frequency signals.  
         DESCRIPTION OF THE RELATED ART  
         [0002]    Heterodyne receivers are frequently used as radio frequency (RF) signal receivers. Heterodyne receivers convert a received RF signal to a fixed intermediate frequency (IF) by mixing, or heterodyning, the received signal with a local signal. By converting received signals to a fixed IF, a heterodyne receiver is able to use fixed-tuned amplifiers and filters, which generally have better selectivity and sensitivity than tunable amplifiers and filters.  
           [0003]    In a superheterodyne receiver, the IF frequency is chosen to be higher than the desired output signal frequency. A simple superheterodyne receiver mixes an incoming RF signal with the output of a local oscillator to produce a fixed intermediate frequency signal. The local oscillator frequency can be adjusted as the input signal frequency changes so as to always produce an intermediate frequency at the same frequency. The mixer output actually consists of two components: an undesired component at a frequency equal to the sum of the input frequency and the oscillator frequency (called the image frequency) and the desired component at a frequency equal to the difference of the input frequency and the oscillator frequency. The image frequency differs from the desired frequency by twice the intermediate frequency and the undesired image frequency component is typically filtered out. In some receivers concerned with receiving higher frequencies, the signal may be mixed in several stages and thus being translated to several different fixed intermediated frequencies before finally being converted to the desired output signal frequency.  
           [0004]    A heterodyne or superheterodyne receiver might be used in a block downconverter. A block downconverter may be configured to receive an RF input and to convert certain blocks or bands of the RF input signal to IF blocks that are centered around a different frequency but have the same bandwidth as the RF band. Block downconverters may be used in a variety of applications, including communication signal analyzers.  
         SUMMARY OF THE INVENTION  
         [0005]    Several embodiments of a block downconverter using a SBAR bandpass filter in a superheterodyne receiver are disclosed. In one embodiment, an apparatus that includes a block downconverter is disclosed. In some embodiments, the apparatus may be a communications signal analyzer. The block downconverter is coupled to receive a radio frequency input. The radio frequency input includes a target region. The block downconverter is configured to produce a selected one of an overlapping plurality of portions of the target region as an intermediate frequency (IF) block having a fixed center frequency. Furthermore, the block downconverter includes a semiconductor bulk acoustic resonator (SBAR) filter that operates as an IF filter in the block downconverter. The SBAR filter may include one or more piezoelectric resonators. In some embodiments, the SBAR filter may include a layer of piezoelectric material, a pair of electrodes mounted on one surface of the piezoelectric material, and a third electrode mounted on an opposing surface of the piezoelectric material so that each electrode of the pair is mounted in overlapping relation to the third electrode to create two series connected resonators that are the only connections to the third electrode. In some embodiments, the two series connected resonators may have identical resonant frequencies. In one embodiment, the SBAR filter may include a piezoelectric resonator-based T network. In another embodiment, the SBAR filter may include a piezoelectric resonator-based pi network. The SBAR filter may include a piezoelectric resonator-based L network in one embodiment.  
           [0006]    In one embodiment, the block downconverter may include a first IF section that is configured to produce a first IF signal having a center frequency of 3.2 GHz. This first IF section may include a SBAR filter that has a center frequency of 3.2 GHz and is configured to filter the first IF signal.  
           [0007]    In another embodiment, a block downconverter is disclosed. The block downconverted includes a radio frequency section coupled to receive a radio frequency input and configured to produce a target region of the radio frequency input. The block downconverter also includes a local oscillator configured to produce a local oscillator signal having a frequency greater than a highest frequency of the target region of the radio frequency input. The block downconverter includes an intermediate frequency (IF) section coupled to receive the target region and the local oscillator signal and configured to heterodyne electromagnetic waves in the target region within the local oscillator signal to produce an IF frequency band, and wherein a lowest frequency of the IF frequency band is greater than the highest frequency of the target region of the radio frequency input. The IF section includes one or more semiconductor bulk acoustic resonator (SBAR) bandpass filters that include at least one SBAR, and at least one SBAR bandpass filter has a center frequency which is the same as a center frequency of the IF frequency band. The target region of the radio frequency input may extend from about 9 kHz to approximately 2.6 GHz. The local oscillator signal may vary from about 3.2 GHz to approximately 5.8 GHz and may be variable in increments of about 1 MHz. The IF frequency band may have a center frequency of about 3.2 GHz.  
           [0008]    In another embodiment, a method of heterodyning a RF signal having a frequency from about 9 kHz to approximately 2.6 GHz is disclosed. The RF signal is received. A 20 MHz band of the received RF signal is mixed with a signal from a local oscillator to produce a first IF band having a center frequency of 3.2 GHz. The first IF band is passed through an SBAR bandpass filter having a center frequency of 3.2 GHz. In some embodiments, the SBAR filter may include a piezoelectric resonator-based T network. In other embodiments, the SBAR filter may include a piezoelectric resonator-based pi network. In one embodiment, the SBAR filter may include a piezoelectric resonator-based L network.  
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0009]    A better understanding of the present invention can be obtained when the following detailed description of the preferred embodiment is considered in conjunction with the following drawings, in which:  
         [0010]    [0010]FIG. 1 illustrates one embodiment of a communications signal analyzer;  
         [0011]    [0011]FIG. 2 illustrates one embodiment of the operation of a block downconverter;  
         [0012]    [0012]FIG. 3 shows one embodiment of a block downconverter;  
         [0013]    [0013]FIG. 4 shows one embodiment of an RF section;  
         [0014]    [0014]FIG. 5 illustrates one embodiment of a first IF section;  
         [0015]    [0015]FIG. 6 a  shows one example of a piezoelectric resonator-based T network;  
         [0016]    [0016]FIG. 6 b  shows one example of a piezoelectric resonator-based pi network;  
         [0017]    [0017]FIG. 6 d  shows one example of a piezoelectric resonator-based L network;  
         [0018]    [0018]FIG. 7 shows one embodiment of a first local oscillator;  
         [0019]    [0019]FIG. 8 illustrates one embodiment of a second IF section;  
         [0020]    [0020]FIG. 9 shows another embodiment of second local oscillator; and  
         [0021]    [0021]FIG. 10 shows one embodiment of a third IF section.  
         [0022]    While the invention is susceptible to various modifications and alternative forms, specific embodiments thereof are shown by way of example in the drawings and will herein be described in detail. It should be understood, however, that the drawings and detailed description thereto are not intended to limit the invention to the particular form disclosed, but on the contrary, the intention is to cover all modifications, equivalents and alternatives falling within the spirit and scope of the present invention as defined by the appended claims.  
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT  
       [0023]    Incorporation by Reference  
         [0024]    The following are hereby incorporated by reference as though fully and completely set forth herein:  
         [0025]    U.S. Pat. No. 5,231,327 titled “Optimized Piezoelectric Resonator-Based Networks, issued to Ketcham; and  
         [0026]    U.S. Pat. No. 5,404,628 titled “Method for Optimizing Piezoelectric Resonator-Based Networks,” issued to Ketcham.  
         [0027]    Description of Figures  
         [0028]    [0028]FIG. 1 is a diagram of one embodiment of a communications signal analyzer (CSA)  10 . CSA  10  includes a block downconverter  12 , an intermediate frequency (IF) digitizer  14 , and a computer system  16 . Block downconverter  12  receives a radio frequency (RF) input. The RF input includes electromagnetic waves, and may include a portion ranging from 9 kHz to 2.6 GHz. One or more RF signals may exist in the portion of the RF input ranging from 9 kHz to 2.6 GHz.  
         [0029]    [0029]FIG. 2 is a diagram illustrating the operation of block downconverter  12 . Block downconverter  12  works as a superheterodyne receiver by converting a received RF signal to a fixed intermediate frequency. Block downconverter  12  converts electromagnetic waves within a selected 20 MHz band or “block” of the portion of the RF input ranging from 9 kHz to 2.6 GHz to an IF frequency band or “block” having a frequency range extending from 5 to 25 MHz, and having a center frequency of 15 MHz. As illustrated in FIG. 2, the 20 MHZ blocks produced by block downconverter  12  are separated by 1 MHz “steps,” and adjacent 20 MHz blocks have a 19 MHz overlap. Block downconverter  12  produces a selected one of an overlapping set of 20 MHz blocks of the portion of the RF input ranging from 9 kHz to 2.6 GHz as a 20 MHz IF frequency block having a fixed center frequency of 15 MHz.  
         [0030]    Referring back to FIG. 1, IF digitizer  14  receives the 20 MHz IF frequency block produced by block downconverter  12 . IF digitizer  14  includes an analog-to-digital converter (ADC) which quantizes and samples the electromagnetic waves present in the IF frequency block, producing digital data indicative of the voltage levels of the electromagnetic waves present in the IF frequency block. IF digitizer  14  may also include circuitry to perform signal processing and/or analysis operations upon the digital data (e.g., filtering, amplification, attenuation, level shifting, Fourier transformation, etc.). IF digitizer  14  provides digital data to computer system  16  derived from the electromagnetic waves present in the IF frequency block.  
         [0031]    Computer system  16  includes a memory  18 , a display device  20 , and an optional printer  22 . Computer system  16  receives the digital data produced by IF digitizer  14 , and stores the digital data in memory  18 . Computer system  16  may include circuitry to perform signal processing and/or analysis operations upon the data (e.g., filtering, amplification, attenuation, level shifting, Fourier transformation, etc.). In response to user input, computer system  16  may display digital data derived from the electromagnetic waves present in the IF frequency block upon display device  20 . The user may also use optional printer  22  to obtain a hard copy of the digital data derived from the electromagnetic waves present in the IF frequency block.  
         [0032]    [0032]FIG. 3 is a diagram of one embodiment of block downconverter  12  of FIG. 1. In the embodiment of FIG. 3, block downconverter  12  includes an RF section  30 , a first IF section  32 , a first local oscillator (LO)  34 , a second IF section  36 , a second LO  38 A, a third IF section  40 , and a third LO  38 B. RF section  30  receives the RF input and produces the portion of the RF input ranging from 9 kHz to 2.6 GHz. IF section  32  receives the portion of the RF input ranging from 9 kHz to 2.6 GHz from RF section  30  and a signal from LO  34 , and produces a first IF band having a center frequency of 3.2 GHz. IF section  36  receives the first IF frequency from IF section  32  and a signal from LO  38 A, and produces a second IF band having a center frequency of 320 MHz. IF section  40  receives the second IF frequency from IF section  36  and a signal from LO  38 B, and produces a third IF band. The third IF band is centered at 15 MHz and extends from 5 to 25 MHz. The third IF band is the 20 MHz IF frequency block produced by block downconverter  12 , and is the selected 20 MHz block of the portion of the RF input ranging from 9 kHz to 2.6 GHz.  
         [0033]    [0033]FIG. 4 is a diagram of one embodiment of RF section  30  of FIG. 3. In the embodiment of FIG. 4, RF section  30  includes an alternating current (AC) coupling network  50 , three switchable attenuators  52 A- 52 C, and a low pass filter (LPF)  54 , all connected in series as shown in FIG. 4. AC coupling network  50  receives the RF input and blocks any direct current (DC) in the RF input. Switchable attenuators  52 A and  52 B receive separate control signals, and each provides either 0 decibels (dB) or 20 dB of attenuation dependent upon the respective control signal. Switchable attenuator  52 C receives a control signal and provides either 0 dB or 10 dB of attenuation dependent upon the control signal. LPF  54  is a filter having a−3 dB corner frequency of 2.6 GHz. LPF  54  produces the portion of the RF input ranging from 9 kHz to 2.6 GHz.  
         [0034]    [0034]FIG. 5 is a diagram of one embodiment of IF section  32  of FIG. 3. In the embodiment of FIG. 5, IF section  32  includes five impedance matching networks  60 A- 60 E, a mixer  62 , two bandpass filters (BPFs)  64 A- 64 B, and two amplifiers  66 A- 66 B, all connected in series as shown in FIG. 5. Impedance matching networks  60 A- 60 E provide needed impedance matching within IF section  32 . Mixer  62  is coupled to receive the portion of the RF input ranging from 9 kHz to 2.6 GHz from RF section  30  and a signal from LO  34 . The signal from LO  34  is variable from 3.2 GHz to 5.8 GHz in increments of about 1 MHz, and the frequency of the signal from LO  34  is selected such that block downconverter  12  produces a desired 20 MHz block of the portion of the RF input ranging from 9 kHz to 2.6 GHz. Mixer  62  heterodynes or mixes the portion of the RF input ranging from 9 kHz to 2.6 GHz with the signal from LO  34 , producing an RF spectrum including a first IF band centered at 3.2 GHz. Having a high first IF improves image rejection.  
         [0035]    BPFs  64 A- 64 B are coupled in series between an output of mixer  62  and an output of IF section  32 . BPFs  64 A- 64 B have center frequencies of about 3.2 GHz and−3 dB bandwidths. BPFs  64 A- 64 B pass the first IF band centered at 3.2 GHz and sufficiently attenuate components of the RF spectrum produced by mixer  62  outside of the bandwidths of BPFs  64 A- 64 B. IF amplifier  66 A is coupled between BPF  64 A and BPF  64 B, and amplifies the first IF band after the first IF band passes through BPF  64 A and before the first IF band passes through BPF  64 B. IF amplifier  66 B is coupled between an output of BPF  64 B and the output of IF section  32 , and amplifies the first IF band after having passed through BPF  64 B.  
         [0036]    BPFs  64 A- 64 B preferably include multiple semiconductor bulk acoustic resonators (SBARs) connected to form an SBAR bandpass filter. SBAR bandpass filters are advantageously smaller than other known types of filters. Suitable SBAR bandpass filters may be obtained from TFR Technologies, Inc., Bend, Oreg. Applicants note that the SBAR bandpass filters from TFR Technologies are advantageously free of resonances over a fundamental. In some embodiments, the SBAR bandpass filters may be similar to those disclosed by U.S. Pat. No. 5,231,327, titled “Optimized Piezoelectric Resonator-Based Networks,” issued to Ketcham or those disclosed in U.S. Pat. No. 5,382,930, titled “Monolithic Multipole Filters Made of Thin Film Stacked Crystal Filters,” issued to Stokes, et al. For example, the SBAR bandpass filters may be include a layer of piezoelectric material, a pair of electrodes mounted on one surface of the piezoelectric material, and a third electrode mounted on an opposing surface of the piezoelectric material so that each electrode of the pair is mounted in overlapping relation to the third electrode to create two series connected resonators that are the only connections to the third electrode. In other embodiments, other suitable SBAR bandpass filters may be used.  
         [0037]    In one embodiment, the SBAR filter may include a piezoelectric resonator-based T network. FIG. 6 a  shows an example of an electrical circuit that includes a piezoelectric resonator-based T network. The T network includes resonator X 1 , series resonator X 2  and shunt element resonator X 3 . In other embodiments, the SBAR filter may include a piezoelectric resonator-based pi network, such as the one exemplified in FIG. 6 b.  In FIG. 6 b,  the pi network includes several series connected resonators X 1 , X 2 , and X 3 . In still other embodiments, the SBAR filter may include a piezoelectric resonator-based L network. FIG. 6 c  shows an example of a piezoelectric resonator-based L network, which includes a series resonator X 1  and a shunt element resonator X 2 .  
         [0038]    [0038]FIG. 7 is a diagram of one embodiment of local oscillator (LO)  34  of FIGS. 3 and 5. In the embodiment of FIG. 7, LO  34  includes a microstrip coupler  70 , an impedance matching network  72 , an amplifier  74 , a prescaler  76 , a phase-locked loop (PLL)  78 , a loop filter  80 , a driver  82 , a digital-to-analog converter (DAC)  84 , and an oscillator  86 . As described above, LO  34  provides a signal to mixer  62  of IF section  32  which is variable from 3.2 GHz to 5.8 GHz in increments of about 1 MHz. The frequency of the output signal of LO  34  is selected such that block downconverter  12  produces a desired 20 MHz block of the portion of the RF input ranging from 9 kHz to 2.6 GHz.  
         [0039]    Oscillator  86  produces the output signal of LO  34  dependent upon a control signal produced by driver  82 . Microstrip coupler  70  is coupled to receive the signal produced by LO  34 , and provides the signal to amplifier  74  via impedance matching network  72 . Amplifier  74  amplifies the signal, and provides the signal to prescaler  76 . Prescaler  76  divides the frequency of the signal by a factor of 2, and provides the resulting prescaled signal to PLL  78 . PLL  78  also receives a 10 MHz clock signal. PLL  78  produces an output signal dependent upon a phase difference between the prescaled signal and the 10 MHz clock signal. Loop filter  80  receives the output signal produced by PLL  78  and filters the output signal. Driver  82  receives the filter output of PLL  78  and an output of DAC  84 . The output of DAC  84  is dependent upon a digital input value. The digital input value is selected by the user in order to select the frequency of the output signal produced by LO  34 . The digital value thus selects the desired 20 MHz block of the portion of the RF input ranging from 9 kHz to 2.6 GHz produced by block downconverter  12 . Driver  82  produces the control signal dependent upon the filtered output of PLL  78  and the output of DAC  84 .  
         [0040]    Oscillator  86  is preferably a current-controlled yttrium iron garnet (YIG) oscillator, and the control signal produced by driver  82  is preferably a current signal.  
         [0041]    [0041]FIG. 8 is a diagram of one embodiment of second IF section  36  of FIG. 3. In the embodiment of FIG. 8, IF section  36  includes a mixer  90 , three amplifiers  92 A- 92 C, two bandpass filters (BPFs)  94 A- 94 B, and two impedance matching networks  96 A- 96 B, all connected in series as shown in FIG. 8. Mixer  90  is coupled to receive the first IF band centered at 3.2 GHz from first IF section  32  and a signal from LO  38 A. The signal from LO  38 A is fixed at 2.88 GHz such that second IF section  36  produces a desired second IF band centered at 320 MHz. Mixer  90  heterodynes or mixes the first IF band centered at 3.2 GHz with the signal from LO  38 A, producing an RF spectrum including the desired second IF band centered at 320 MHz. IF amplifier  92 A is coupled between an output of mixer  90  and an input of BPF  94 A, and amplifies the second IF band centered at 320 MHz before the second IF band is passed through BPF  94 A.  
         [0042]    BPFs  94 A- 94 B are coupled in series between the output of mixer  90  and an output of IF section  36 . BPFs  94 A- 94 B have center frequencies of about 320 MHz and −3 dB bandwidths of about 22 MHz. BPFs  94 A- 94 B pass the second IF band centered at 320 MHz and sufficiently attenuate all components of the RF spectrum produced by mixer  90  outside of the 22 MHz bandwidth of BPFs  94 A- 94 B. Impedance matching networks  96 A- 96 B provide needed impedance matching within IF section  36 . IF amplifier  92 B is coupled between BPF  94 A and BPF  94 B, and amplifies the second IF band centered at 320 MHz after the second IF band is passed through BPF  94 A and before the second IF band is passed through BPF  94 B. IF amplifier  92 C is coupled between an output of BPF  94 B and the output of IF section  36 , and amplifies the second IF band centered at 320 MHz after having passed through BPF  94 B.  
         [0043]    BPFs  94 A- 94 B are preferably include multiple surface acoustic wave (SAW) resonators connected to from a SAW bandpass filter. Suitable SAW bandpass filters are available from Sawtek Incorporated in Orlando, Fla.  
         [0044]    [0044]FIG. 9 is a diagram of one embodiment of local oscillator (LO)  38  representative of LO  38 A of FIGS. 3 and 8 and LO  38 B of FIGS. 3 and 10. In the embodiment of FIG. 9, LO  38  includes a phase-locked loop (PLL)  100 , a loop filter  102 , and an oscillator  104 . As described above, LO  38 A provides a signal to mixer  90  of second IF section  36  which is fixed at 2.88 GHz such that second IF section  36  produces the desired second IF band centered at 320 MHz. As will be described below, LO  38 B provides a signal to a mixer of third IF section  40  which is fixed at 335 MHz such that third IF section  40  produces a desired third IF band centered at 15 MHz.  
         [0045]    Oscillator  104  produces an output signal FOUT of LO  38  dependent upon a control signal produced by loop filter  102 . Oscillator  104  is preferably a voltage-controlled oscillator, and the control signal produced by loop filter  102  is preferably a voltage signal. PLL  100  receives the output signal and the 10 MHz clock signal. PLL  100  produces an output signal dependent upon a phase difference between the output signal and the 10 MHz clock signal. Loop filter  102  receives the output signal produced by PLL  100  and filters the output signal to produce the control signal.  
         [0046]    [0046]FIG. 10 is a diagram of one embodiment of third IF section  40  of FIG. 3. In the embodiment of FIG. 10, IF section  40  includes a mixer  110 , a first impedance matching network  112 , a switchable attenuator  114 , a bandpass filter (BPF)  116 , two amplifiers  118 A- 118 B, and a second impedance matching network  120 , all connected in series as shown in FIG. 10. Mixer  110  is coupled to receive the second IF band centered at 320 MHz from second IF section  36  and a signal from LO  38 B. The signal from LO  38 B is fixed at 335 MHz such that third IF section  40  produces a desired third IF band centered at 15 MHz.  
         [0047]    Mixer  110  heterodynes or mixes the second IF band centered at 320 MHz with the signal from LO  38 B, producing an RF spectrum including the desired third IF band centered at 15 MHz. Impedance matching networks  112  and  120  provide needed impedance matching within IF section  40 . Switchable attenuator  114  receives a control signal and provides either 0 dB or 10 dB of attenuation dependent upon the control signal. BPF  116  is coupled in series between an output of mixer  110  and an output of IF section  40 . BPF  116  has a center frequency of about 15 MHz and −3 dB corner frequencies of approximately 1 MHz and 50 MHz. BPF  116  passes the third IF band centered at 15 MHz and sufficiently attenuates all components of the RF spectrum produced by mixer  110  above and below the −3 dB corner frequencies of BPF  116 . IF amplifiers  118 A- 118 B are coupled between an output of BPF  116  and an output of IF section  40 , and amplify the third IF band centered at 15 MHz after having passed through BPF  116 .  
         [0048]    Block downconverter  12  produces the third IF band produced by IF section  40 . The third IF band has a −3 dB bandwidth of about 20 MHz. As described above, the third IF band is the 20 MHz IF frequency block produced by block downconverter  12 , and is the desired 20 MHz block of the portion of the RF input ranging from 9 kHz to 2.6 GHz.  
         [0049]    Numerous variations and modifications will become apparent to those skilled in the art once the above disclosure is fully appreciated. It is intended that the following claims be interpreted to embrace all such variations and modifications.