Abstract:
In frequency channel communications, constraints on transmit power characteristics can be complied with by implementing constellation shaping ( 15, 71 ) in the transmitter. In transmitters which use transmit filtering ( 91 ) to comply with constraints on transmit power characteristics, the transmitter can also implement, upstream of the transmit filter, precoding ( 92 ) designed for a desired cooperation with the transmit filter.

Description:
This application claims the priority under 35 USC 119(e)(1) of U.S. provisional application Nos. 60/343,651 and 60/344,154, both filed on Dec. 28, 2001 now abandoned and both incorporated herein by reference. 

   FIELD OF THE INVENTION 
   The invention relates generally to frequency channel communications and, more particularly, to spectral power management in frequency channel communications. 
   BACKGROUND OF THE INVENTION 
   In various fields of communications, certain spectral requirements are imposed on transmitters due, for example, to regulatory limitations and/or interference considerations. In Home Phoneline networking (HomePNA), for example, these include: (1) FCC regulations, specifically part 15 (radiated emissions) and part 68 (conducted emissions), wherein the latter requires averaging the power over a period (time window) of 2 uS; (2) avoidance of audible noise in POTS (when apparent), which imposes peak constraints that seem to match the ones of part 68; and (3) avoidance of interference with HAM RF, wherein it is assumed that transmitting below −80 dBm/Hz in the HAM bands is sufficient. 
   Compatibility with FCC part 15 and 68 imposes constraints on the peak power, whereas the requirement to avoid interfering with HAM bands imposes requirements on the transmitted power in HAM bands (e.g. around 7 MHz). In the HomePNA2.0 specification, the foregoing requirements were met by the following solutions: using a PSD mask that complies with the FCC regulations and the HAM RF egress restrictions; normalizing the transmitted signal constellation according to the peak power (outermost symbols) to follow peak constraints; and using notch filters in the transmitter to comply with the power restrictions in the HAM bands. Normalizing the signal constellation according to the outermost symbols can cause a loss of up to 5 dB in large constellations. In the 256QAM constellation used in HPNA2.0, this imposes a loss of 4.23 dB. Using notch filters in the transmitter can result in a transmitted pulse that suffers from ISI and a longer impulse response, which in turn might degrade noise performance, and enhance error propagation in the receiver. 
   It is therefore desirable to provide spectral power management schemes that avoid undesired effects such as described above. 
   The invention attempts to avoid such undesired effects by implementing constellation shaping in the transmitter to support compliance with constraints on transmit power characteristics such as transmit power during a time window and/or transmit power in one or more predetermined frequency bands. Some embodiments of the invention provide preceding in the transmitter to avoid undesirable effects that can occur when special transmit filtering is used in the transmitter to comply with constraints such as constraints on transmit power in one or more frequency bands. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  diagrammatically illustrates pertinent portions of exemplary embodiments of a transmitter for frequency channel communications according to the invention. 
       FIG. 2  diagrammatically illustrates pertinent portions of a conventional receiver apparatus which is capable of communicating with the transmitter of  FIG. 1 . 
       FIG. 3  diagrammatically illustrates exemplary embodiments of the metric element of  FIG. 1 . 
       FIG. 4  diagrammatically illustrates further exemplary embodiments of the metric element of  FIG. 1 . 
       FIG. 5  diagrammatically illustrates further exemplary embodiments of the metric element of  FIG. 1 . 
       FIG. 6  diagrammatically illustrates further exemplary embodiments of the metric element of  FIG. 1 . 
       FIG. 7  diagrammatically illustrates pertinent portions of exemplary embodiments of a transmitter for frequency channel communications according to the invention. 
       FIG. 8  diagrammatically illustrates pertinent portions of exemplary embodiments of a transmitter for frequency channel communications according to the invention. 
       FIG. 9  diagrammatically illustrates pertinent portions of exemplary embodiments of a transmitter for frequency channel communications according to the invention. 
   

   DETAILED DESCRIPTION 
   The documents listed below are all incorporated by reference herein. Each document is hereinafter referred to by the corresponding number shown below in square brackets to the left of the document. 
   [1] “Trellis shaping,” G. D. Forney IEEE Trans. Inf., Vol. 38, March 1992. 
   [2] “Trellis Precoding: combined coding, preceding and shaping for intersymbol interference channels,” M. V. Eyuboglu, G. D. Forney, IEEE Trans. Inf., Vol. 38, March 1992. 
   [3] “On optimal shaping of multidimensional constellations,” R. Laroia, N. Farvardin, S. A. Tretter, IEEE Trans. Inf., Vol. 40, July 1994. 
   [4] “New automatic equalizer employing modulo arithmetic,” M. Tomlinson, Electron. Lett., Vol. 7, pp. 138–139, March 1971. 
   [5] “Trellis Precoding: combined coding, preceding and shaping for intersymbol interference channels,” M. V. Eyuboglu, G. D. Forney, IEEE Trans. Inf., Vol. 38, March 1992. 
   [6] “A simple and effective precoding scheme for noise whitening on intersymbol interference channels,” R. Laroia, S. A. Tretter, N. Farvardin, IEEE Trans. Commun., October 1993. 
   [7] ITU-T Recommendation, V.34, September 1992. 
   [8] “More on Convolutional Spectral Shaping,” V. Eyuboglu, V.pcm Rapporteur Meeting, La Jolla, Calif., May 5–7, 1997. 
   [9] ITU-T Recommendation, V.90, September 1998. 
   FCC regulation part 68, for example, imposes a restriction on the transmitted power, when averaged on a 2 microsecond window. When the transmitted symbol rate is low, this might impose a restriction on each transmitted symbol, but as symbol rates increase, it is possible to select symbols such that, in each 2 microsecond window, the average power complies with the regulation. This may be done by using shaping (e.g. a variant of trellis shaping as in [1] or Convolutional spectral shaping as in [8]). The shaping operation permits the use of symbols that were not allowed by other methods (e.g. single symbol peak power constraints), which in turn results in better performance (e.g. higher throughput or better robustness to noise). 
   As indicated above, power constraints are often imposed in frequency bands that are close to or even within the frequency band of the desired transmission. For example, HAM bands in the 7 MHz range are within the 4–10 MHz frequency band used by HomePNA2.0. This problem arises also for VDSL. The aforementioned notch filter solution may introduce undesired inter-symbol interference (ISI), which in turn might degrade performance. 
   According to some embodiments of the invention, precoding (e.g. Tomlinson precoding as in [4] or Laroia preceding as in [6]) may be used before the notch filter to generate a spectrally shaped, non-ISI signal at the transmitter output. Other embodiments use a shaping technique (e.g. a variant of trellis shaping as in [1] or convolutional spectral shaping as in [8]) that imposes a spectral constraint on the designated HAM bands. In this manner, a sequence of transmitted symbols can be selected such that the power in the HAM bands is minimized. 
   Constellation shaping permits the stream of transmitted symbols in a communication system to be selected according to a criterion of minimum average power under the constraint of a given minimum distance between neighboring points. This “shapes” the constituent 2-dimensional constellation into a certain form (with a certain probability distribution between the constellation points). Thus, a symbol sequence with a lower average power (or alternatively a symbol sequence with a higher inter-symbol distance between neighboring points for a given power constraint) can be used. This results in “shaping gain” which can make the communication system more robust to noise and channel impairments, or result in higher achievable data-rates. 
   Two known methods of constellation shaping are trellis shaping as in [1] and shell mapping as in [3]. On Gaussian channels with inter-symbol interference (ISI), it is often desired to use preceding methods to mitigate channel distortion. Tomlinson-Harashima (TH) preceding as in [4] is a well known preceding scheme for ISI channels. 
   When constellation shaping is used in ISI channels, it can be advantageous to use a combination of shaping and preceding. A method of combining trellis shaping with TH-preceding, called trellis preceding, is described in [3]. Combining shell mapping with precoding can be done by incorporating Laroia preceding as described in [6]. In [1] and [2] it is shown that lattice codes can also be combined with trellis shaping and trellis preceding. Thus, trellis coded modulation (TCM) schemes can be combined with shaping and preceding to achieve coding gain together with shaping gain, and to have high performance even in ISI channels. 
   Lattice codes (e.g. TCM) can also be combined with shell mapping and Laroia preceding to achieve coding gain together with shaping gain, and to have high performance even in ISI channels. This is done, for example, in the ITU V.34 standard (see [7]) for voice grade modems. 
   In the ITU V.90 standard for voice grade modems (see [9]), convolutional spectral shaping as in [8] is used to spectrally shape the transmitted signal, or in other words, minimize the transmitted energy in predefined frequency bands. Notice that in the case of V.90 telephony modems, this band is the DC band (the requirement for minimizing the transmitter power results from the existence of transformers that stop the very low frequencies). 
     FIG. 1  and  FIG. 2  respectively illustrate pertinent portions of exemplary embodiments of a transmitter and receiver for using trellis shaping for spectral management. Referring to  FIG. 1 , the input sequence is divided into three parts (x j , w j , s j ). The first part, x j , a binary k c -tuple, is an input to an encoder for a rate k c /n c  TCM code (or other type of lattice code). The second part, w j , is an uncoded binary n u  tuple. The third part, s j , a syndrome r s -tuple, is an input to an r s  input, n s  output coset representative generator (H s   −1 ) T  for a rate k s /n s  convolutional shaping code, where k s =n s −r s . The signals t j , w j  and y j  are input to the decoder  15 , whose output y s,j  is summed with t j  to produce z j . The signals w j , y j  and z j  are input to a symbol mapper. Except for the design of the metric element  17 , the transmitter of  FIG. 1  can have a conventional design, for example, generally following section III(A) of [1]. The decoder element  15  can, in conventional fashion, use the metric information output by the metric element  17 . Thus, when the decoder element is implemented using a Viterbi algorithm (VA), the metric element is implemented per each branch. The design of the metric element can vary according to the desired spectral management criteria. 
   Referring to  FIG. 2 , and with the exception of the broken line portion (discussed in more detail hereinbelow), the illustrated receiver of  FIG. 2  is conventional, and generally follows section III(C) of [1]. This receiver is operable in conventional fashion to receive (e.g. via conventional phone lines) communications from the transmitter of  FIG. 1  (and the transmitters of  FIGS. 7–9  below). 
   For complying with restrictions on transmitted power over a time-window,  FIG. 3  shows exemplary embodiments of the metric element  17  of  FIG. 1 . The magnitude squaring element  31  squares the magnitude of the signal point a j  produced by the symbol mapper of  FIG. 1 , and thus calculates a measure of the transmit signal power. The averaging filter  33  may be given by: 
             avg   j     =       1   N     ⁢       ∑     i   =   0       N   -   1       ⁢           ⁢            a     j   -   i            2               
where N is the number of symbols used for the averaging function, and can be set according to the ratio between the time-window for the power constraint and the symbol interval. For example, if the symbol rate is 4 Mbaud, compliance with FCC part  68  (2 uSec window) yields N=8. The output of metric function  35  may be given by:
 
             m   j     =     {         ∞           avg   j     &gt;   Threshold                       ⁢            a   j          2           otherwise                 
where the Threshold value is set according to the power constraint.
 
   The above example allows for trellis shaping, without permitting sequences of N consecutive symbols to have an average power greater than the predefined threshold. Notice that setting the metric function output to infinity is equivalent to disconnecting certain branches in the Viterbi algorithm (VA) implemented by the decoder  15  of  FIG. 1 . Further notice that for an averaging filter of length N, each state of the convolutional code associated with the VA of decoder  15  should be theoretically partitioned into D (N−1)  states (D being the constellation size), according to all possible combinations of the last N−1 symbols (that yield a different value for the filter output). This can in turn yield a complex VA at  15  in  FIG. 1 . 
   To avoid such a complex VA, it is possible to use conventional reduced state sequence estimation (RSSE), for example parallel decision feedback decoding (PDFD, see [2]), i.e. to attach a shift register to every state of the decoder  15 , each shift register holding the last N−1 symbol decisions associated with the corresponding state. Using these shift registers, the averaging filter output can be calculated, and negligible complexity enhancement is needed. Such a PDFD embodiment is shown by broken line in  FIG. 3 . 
   In some embodiments according to  FIG. 3 , a trellis shaped symbol sequence is obtained using 4 information bits per symbol (i.e., using a shaping constellation of 32-QAM), and an averaging filter with N=8. 
   For complying with restrictions on transmit power over a frequency band, an exemplary embodiment of metric element  17  is given in  FIG. 4 . The output of the band pass filter  41  may be given by: 
             BPF   j     =         ∑     i   =   0       L   -   1       ⁢           ⁢       b   i     ⁢     a     j   -   i           -       ∑     i   =   1       K   -   1       ⁢           ⁢       d   i     ⁢     BPF     j   -   i                   
where b, d , K and L define the taps of the band pass filter. These tap parameters are set according to the frequency band in which the power constraint applies. The output of metric function  43  may be given by:
   m   j =|BPF j | 2   
   The  FIG. 4  example allows for trellis shaping, wherein the power is calculated only in the frequency band in which the constraint applies. Notice that PDFD may also be used for implementing the BPF, by holding two registers attached to every state of decoder  15 , the two registers of each state respectively holding the last L−1 symbol decisions and the last K−1 band pass filter outputs associated with the corresponding state. Using these shift registers, the BPF output can be calculated, and negligible complexity enhancement is needed. Such a PDFD embodiment is shown by broken line in  FIG. 4 . 
   In some embodiments according to  FIG. 4 , a trellis shaped symbol sequence is obtained using 4 information bits per symbol (i.e. using a shaping constellation of 32-QAM), and a 2 nd  order Butterworth band pass filter. 
   For complying with restrictions on both the transmitted power over a time-window as well as the transmitted power over a frequency band, an exemplary embodiment of metric element  17  is given in  FIG. 5 . The exemplary metric element of  FIG. 5  includes the magnitude squaring element  31  and averaging filter  33  of  FIG. 3 , and the band pass filter  41  of  FIG. 4 . The output of metric function  51  may be given by: 
             m   j     =     {         ∞           avg   j     &gt;   Threshold                   k   1     ⁢           ⁢            a   j          2       +       k   2     ⁢     BPF   j             otherwise                 
where the Threshold value is set according to the power constraint, and the weighted sum coefficients k 1  and k 2  are set according to the desired proportions between shaping according to overall power (see  FIG. 3 ) and shaping where the power is calculated only in the frequency band in which the constraint applies (see  FIG. 4 ). The aforementioned coefficients can be determined, for example, empirically based on experimental observation or simulation under expected operating conditions. The  FIG. 5  example allows for trellis shaping, without permitting sequences of N consecutive symbols to have a power greater than the predefined threshold. Although not explicitly shown, the aforementioned use of PDFD registers is also applicable to the embodiments of  FIG. 5 .
 
   In some embodiments according to  FIG. 5 , a trellis shaped symbol sequence is obtained using 4 information bits per symbol (i.e. using a shaping constellation of 32-QAM), an averaging filter with N=8, a 2 nd  order Butterworth band pass filter, and a metric function with k 1 =0.1 and k 2 =1. 
   Some exemplary embodiments of the metric element  17  of  FIG. 1  can also incorporate preceding (e.g., trellis precoding as in [2], using a TH-precoder). An exemplary metric element that incorporates precoding is shown in  FIG. 6 . The metric element of  FIG. 6  includes a TH precoder  61 , combined with the magnitude squaring element  31  and averaging filter  33  of  FIG. 3 , and the band pass filter  41  of  FIG. 4 . 
   The metric function  63  may be given by: 
             m   j     =     {         ∞           avg   j     &gt;   Threshold                   k   4     ⁢           ⁢            a   j          2       +       k   5     ⁢     BPF   j       +       k   3     ⁢            TH   j          2             otherwise                 
where the Threshold value is set according to the power constraint, wherein the weighted sum coefficients k 4 , k 5  and k 3  are set in a way that gives the desired proportions between (1) shaping according to overall power, (2) shaping where the power is calculated only in the frequency band in which the constraint applies, and (3) precoding, and wherein TH j  is the precoder output. The aforementioned coefficients can be determined, for example, empirically based on experimental observation or simulation under expected operating conditions.
 
   The  FIG. 6  example allows for trellis shaping, without permitting sequences of N consecutive symbols to have a power greater than the predefined threshold. 
   Although not explicitly shown, the aforementioned use of PDFD registers is also applicable to the embodiments of  FIG. 6 , including the TH precoder portion. The use of PDFD with a precoder is described in [2]. As demonstrated in [1], shaping maybe combined with other schemes such as TCM, Turbo-TCM, and RS-coding. 
     FIG. 7  diagrammatically illustrates pertinent portions of exemplary embodiments of a transmitter that uses shaping techniques in the manner described generally above according to the invention. The shaping device, for example decoder element  71 , is controlled by a shaping controller  72  according to a desired criterion. In some embodiments the criterion (or criteria) can be implemented, for example, by a metric element such as one of the exemplary metric elements described above relative to  FIGS. 3–6 . In such embodiments, the metric functions  35 ,  43 ,  51  and  63  serve as control information determiners that determine what control information will be applied to the shaping decoder. Any desired shaping method, for example, trellis shaping (see [1]) or convolutional spectral shaping (see [8] and [9]), can be implemented at  71 . As described above, the criterion (or criteria) of the metric element ensures that the shaping operation at  71  produces shaped information bits at  73  that result (after mapping at  74 ) in a transmitted symbol stream according to the defined requirement(s). 
   As discussed above, when restrictions on the transmitted power over certain frequency bands apply, a special (e.g. notch) filter can be used in the transmitter to attenuate the signal in these bands. Exemplary transmitter embodiments according to the invention can reduce the ISI effect of these filters by using TH preceding upstream of the filter. 
   A block diagram of exemplary transmitter embodiments according to the invention is shown in  FIG. 8 . The ISI introduced by the transmitter filter (e.g., a filter with notches for HAM bands)  81  is dealt with in the transmitter itself, so the equivalent channel seen by the receiver is less severe. Furthermore, this reduces the error propagation phenomenon when the receiver employs a DFE (decision feedback equalizer). The preceding filter  82  can be tailored for use with the known impulse response of the filter  81 . For example, as shown in  FIG. 8 , filter  82  may be an estimate of the inverse (1/h(D)) of filter  81  (h(D)). In contrast, conventional applications of TH precoding typically tailor the precoder filter to the characteristics of the equivalent channel filter for the entire channel between transmitter and receiver. 
   In the transmitter of  FIG. 8 , the transmitter signal will have notches in the desired bands, but ISI can be avoided due to the TH precoder. The TH precoder includes a modulo function at  85  to fold signal points back into the constellation (according to conventional TH precoder operation), so the precoded symbols at  83  are within the selected constellation. A corresponding modulo function would be implemented in conventional fashion at the receiver, as shown by broken line in  FIG. 2 . 
   In other embodiments, different precoding schemes, e.g. Laroia precoding (see [6]), are used. This is shown generally in  FIG. 9 , where preceding is applied at  92 , upstream of a specialized filter  91  in the transmitter. 
   The above-described embodiments may be implemented in many cases where restrictions on the transmitted power in certain frequency bands apply, such as in HomePNA or VDSL. It will be apparent to workers in the art that these embodiments can be readily implemented, for example, by suitable modifications of software, hardware, or both, in conventional transmitters and receivers, such as HomePNA and VDSL transmitters and receivers. 
   Although exemplary embodiments of the invention are described above in detail, this does not limit the scope of the invention, which can be practiced in a variety of embodiments.