Abstract:
A synchronous rectifier circuit, comprising a power transformer which has a primary side, including first and second primary winding sections, and a secondary side, including first and second secondary winding sections, a rectifier circuit at the secondary side of the power transformer, which rectifier circuit comprises first and second MOSFETs associated with the first and second secondary winding sections, respectively, first and second current transforming means associated with the first and second secondary winding sections, respectively, and first and second drive circuits for the first and second MOSFETs, respectively, each current transforming means generating first and second currents which are dependent on the current of the associated secondary winding section of the power transformer, and each drive circuit comprising first and second branches to receive the first and second currents, respectively, generated by the current transforming means, and the first branch comprising a diode and a transconductance choke, and the second branch comprising a diode.

Description:
This application claims priority to the filing date of German Patent Application Nos. 101 52 901.5 filed Oct. 26, 2001, and 102 32424.7 filed Jul. 17, 2002, and PCT Application No. PCT/EP02/11641 filed Oct. 17, 2002; the specification of all of these applications are incorporated herein in their entirety. 
   FIELD OF THE INVENTION 
   The invention relates to a sychronous rectifier circuit for use in a push-pull d.c. voltage converter. 
   BACKGROUND OF THE INVENTION 
   It is the task of d.c. voltage converters to transform a direct voltage applied at their input into a direct voltage which is output at a different value, and to do so most efficiently. The output value may be greater or smaller than the input value and may be adjustable. In a d.c. voltage converter the direct voltage input, first, is transformed with the help of switching stages into alternating voltage having a rectangular waveshape. During the switch-on period “chopped” direct voltage is stored in the form of magnetic energy in a choke. During the switch-off period, it acts as a self-induction voltage at the output of the d.c. voltage converter. D.C. voltage converters operating according to this principle are referred to as choke converters. They have their inputs and outputs galvanically separated. It is known in the art to use transformers to achieve separation of potential. Here, the induced voltage occurs at the secondary winding, and the voltage transformation can be determined by the number of windings.  FIG. 1  is a block diagram illustrating a transformer-type converter used as d.c. voltage converter.  FIG. 1  depicts the basic elements of a transformer-type converter, including an input switching stage  10 , a power transformer  12 , a rectifier circuit  14 , and an output filter  16 . A distinction is made in the art between single phase d.c. converters and push-pull voltage transformers. A single phase d.c. converter may be regarded as being a simple electronically controlled switch, whereas switch-over operations occur with push-pull voltage transformers, and a transformer  12  having two primary windings may be required. Push-pull transformers can be derived from two single phase d.c. converters connected in parallel. The electronic switch-over is accomplished by two switching stages, and current always flows through one of the two primary windings. The invention relates to the field of push-pull voltage transformers. 
   In practice, such push-pull voltage transformers are employed in switch mode power supply, such as server architecture for telecommunications applications, in PCs, industrial applications, and many other situations. The invention is especially advantageous in distributed energy supply systems where several stages are connected in succession. In new server architectures, for instance, power supply units are used in which the mains voltage, to begin with, is converted into a bus voltage of some 48 to 50 V. A second conversion to +12 V, for example, then takes place within the server sub-system. The specific voltages required for the various components, such as the microprocessor, RAMs, etc. are produced locally by so-called voltage regulator modules which are connected to the 12 V rail. 
   Each of the converting stages must operate as efficiently as possible in view of the series connection of different power stages. For maximum efficiency to be obtained with switch mode power supply, optimization in terms of energy loss is required regarding each and every source thereof. Energy losses depend not only on the type of converter or transformer chosen, be it a single phase d.c. converter or a push-pull voltage transformer, but are determined decisively by the mode of operation of the rectifier circuit. 
   Making good use of the inductivities as well as driving the transducer in positive and negative directions present great advantages of push-pull voltage transformers. Another advantage is the great efficiency of the transformer and the high output power attained. 
     FIG. 2  is a schematic circuit diagram of a push-pull voltage transformer provided with Schottky diodes for rectification at the secondary side. The push-pull forward converter illustrated in  FIG. 2  is known; its function is described, for example, in Billings, Keith “Switch Mode Power Supply Handbook”, 2nd edition, McGraw-Hill, New York, 1999. This converter comprises a power transformer  18  having a primary side  18   a  and a secondary side  18   b . The primary side  18   a  and the secondary side  18   b  each comprise two winding sections. For driving purposes, two power transistors  20 ,  22  are associated with the two winding sections at the primary side  18   a . Two secondary diodes  24  and  26  are associated, respectively, with the two winding sections at the secondary side  18   b . These diodes are connected to an output filter stage made up of a storage choke  28  and a storage capacitor  30 , as may be seen in  FIG. 2 . The power transistors  20 ,  22  are driven, for instance, by a control, IC (not shown). 
   As the transistor  22  is driven, current will flow through the associated winding section of the power transformer  18  and also through the transistor  22 . The polarity of the associated winding section at the secondary side  18   b  of the power transistor  18  causes the diode  26  to cut off. At the same time, voltage is induced also in the other winding section at the secondary side  18   b , thus causing current to flow through the diode  24  via the storage choke  28 . When a sufficient amount of energy has been transmitted from the primary side to the secondary side, the transistor  22  blocks. During the next cycle the transistor  20  is driven. The current now flowing through the second winding section of the primary side  18   a  causes reversal of the polarity of the associated winding at the secondary side  18   b . Diode  24  blocks, while diode  26  is conducting, thus permitting current to flow through the choke  28 , as during the first cycle. To make sure that the two power transistors  20 ,  22  will not be conducting at the same time, a compulsory break, the so-called freewheeling phase, is provided between the first and second cycles described above. During this freewheeling phase, the electric circuit at the secondary side  18   b  of the transformer  18  is formed of the storage choke  28 , the storage capacitor  30 , the two conducting diodes  24  and  26 , and the connected load (not shown in the drawing). 
     FIG. 3  presents idealized waveshapes of the output voltages u 01 , U 02  at the two winding sections of the secondary side  18   b  of the power transformer  18 , forward currents i 01 , i 02  through the diodes  24  and  26 , and the output current i 0  through the storage choke  28 . 
   A positive voltage u 01  is generated during the first time interval from t 1  to t 2 . Diode  24  is conducting. The overall output current i 0  is passed through the same and through the upper secondary winding section of the power transformer  18  toward the output. The rise in output current is determined by the voltage difference u 01 −u 02  (output voltage) and the sum of the inductivities of the secondary circuit. 
   The second time interval from t 2  to t 3  corresponds to the so-called freewheeling phase. The output voltages u 01  and u 02  of the transformer  18  are zero. The current i 0  is determined by the inductivities of the secondary circuit. If the upper and lower winding sections at the secondary side  18   b  are identical the output current i 0  is divided in two. Each of the diodes  24 ,  26  will carry one half of the output current i 0 . During this time interval the output current drops. 
   During the third time interval from t 3  to t 4  a positive voltage u 02  is produced and diode  26  is conducting. The resulting behavior corresponds to that of the first time interval. 
   During the last time interval from t 4  to t 5  of period T both power transistors  20 ,  22  are turned off. The voltages u 01  and u 02  once again are zero, which corresponds to the freewheeling phase. 
   In the embodiment of the d.c. voltage converter shown in  FIG. 2  the secondary rectifier is embodied by diodes. The rectifier diodes produce losses which depend on the forward voltage of the diodes  24 ,  26  and are composed of forward losses and switching losses of the diodes. 
   The forward loss P DC  of a diode is given by the product of its forward voltage drop u F  and its forward current i D  (see also FIG.  4 )
 
 P   DC   =u   F   ·i   D .
 
   The forward voltage rises as the load increases; it lies between 0.5 V and 1.5 V, depending on the type of diode provided. If the transducer output voltage is 3.3 V, for example, which would correspond to a processor voltage, as much as 30% of the voltage will drop at the rectifier diodes. With higher transducer output voltages, e.g. 48 V in telecommunications applications, the voltage drop at the diodes is comparatively less, but still not negligible. 
   The switching loss of a diode can be estimated by the following equation:
 
 P   DS   =Q   F   ·û·f 
 
   where Q F  is the recovered load during the fall time of the reverse current of the diode, f is the reciprocal value of period T, and-is the peak value of the diode turnover voltage. 
   A reduction of the forward loss discussed above can be achieved only by reducing the voltage drop. 
   One solution resides in the use of a MOSFET connected in parallel with the diode. That is shown in  FIG. 4 . The MOSFET is turned on when current in forward direction is applied to the diode, and it is turned off when the current is reversed. This is called synchronous rectification. The diode, such as diodes  24 ,  26 , in a circuit may be replaced by a MOSFET. If the MOSFET used is of vertical structure its antiparallel diode or inverse diode (body diode) is utilized. This is illustrated in  FIG. 5  which shows like members identified by like reference numerals as in  FIG. 1 .  FIG. 5  diagrammatically illustrates the replacement of the diode-type rectifier  14  by a synchronous rectifier circuit  32  on the basis of MOSFETs. 
   With reference to  FIGS. 4 and 5  it becomes clear that the voltage drop u DS  at the MOSFET is determined by the switch-on resistance R DS(ON)  of the MOSFET and the actual drain current which must equal the diode current i D . The following must apply if the energy loss is to be reduced:
 
| u   DS   |=|R   DS(ON)   ·i   D   |&gt;u   F .
 
   The forward loss, therefore, can be reduced by selecting a MOSFET of which the forward resistance R DS(ON)  is small. 
   A control signal is required for the MOSFET to be switched on and off. Generating the control signal has a decisive influence on the switching behavior. Moreover, the energy losses in this circuit must be taken into consideration. There are various known methods of driving synchronous rectifiers comprising a MOSFET, and they may be roughly classified as self-controlled, IC controlled, and current controlled. 
     FIG. 6  is a simplified diagram of the secondary side of a self-controlled synchronous rectifier circuit. Again the same reference numerals are used to designate members corresponding to those of  FIG. 2 . The power diodes  24 ,  26  shown in  FIG. 2  are replaced by MOSFETs  34  and  36 , respectively. The first winding section at the secondary side  18   b  is designated L S1 , and the second winding section is designated L S2 . 
   In the case of the self-controlled synchronous rectifier according to  FIG. 5  the output voltage of the power transformer  18  is used to control the MOSFETs  34 ,  36 . This circuit has the advantage of necessitating only little expenditure in circuitry since no additional driver circuits are needed to drive the MOSFETs  34 ,  36 . 
   With reference to  FIG. 3 , the output voltage u 01  at the first winding section L S1  is positive during the time interval from t 1  to t 2 , while u 02  is negative at the second winding section L S2 . With these conditions, the p-channel MOSFET  34  is switched on because its gate voltage is negative. Corresponding switching behavior is true of the p-channel MOSFET  36  during the time interval from t 3  to t 4 ; MOSFET  34  is turned off, while MOSFET  36  is turned on. Operation of the MOSFET switches  34 ,  36  during these two time intervals is satisfactory. During the freewheeling phase, however, no control voltage is generated. The current flows through the inverse diodes of the MOSFETs, whereby higher forward losses are produced than necessary. 
   Furthermore, it is disadvantageous with the self-controlled synchronous rectifier design that p-channel MOSFETs are needed which are much more expensive and have a higher forward resistance than comparable n-channel MOSFETs. Moreover, the transformer output voltage range is restricted due to the gate voltage of the MOSFETs  34 ,  36 . It must be higher than the threshold voltage and lower than the maximum permissible gate voltage of the MOSFETs of approximately 30 V. 
     FIG. 7  is a circuit diagram of the secondary side of a synchronous rectifier with IC drive, corresponding members being designated by the same reference numerals as in  FIG. 6 . With this synchronous rectifier design, driver ICs  38 ,  40  are provided to drive the MOSFETs  34 ,  36 . There are only a few manufacturers who offer such specific driver ICs for synchronous rectifiers. The IC components  38 ,  40  scan the secondary voltages of the transformer  18 , and the MOSFETs  34 ,  36  are turned on or off, depending on the potential profile. The electronic control assures synchronous switching on and off of the synchronous rectifier. However, the scarce commercial availability and the relatively high costs and greater expenditure involved in connecting and feeding the driver ICs  38 ,  40  are points against making use of driver ICs  38 ,  40 . 
     FIG. 8 , finally, is a circuit diagram of the secondary side of a current controlled synchronous rectifier, showing only the upper part of the secondary side  18   b  which includes the first secondary winding section L S1 . The structure of the lower part, including the second secondary winding section L S2  is mirror inverted. 
   In the current controlled synchronous rectifier, the power MOSFET  34  (and  36 , too, not shown in  FIG. 8 ) is controlled through a current transformer  42 . The current transformer  42  is connected in series between the upper winding section L S1 , at the secondary side  18   b  of the power transformer  18  and the MOSFET  34  and comprises a primary winding  42   a  and a secondary winding  42   b . The secondary winding  42   b  is connected to the gate of the MOSFET  34  by way of a voltage divider composed of two resistors  44 ,  46 . 
   When the transformer  18  is controlled such that current flows through the winding section L S1 , of the secondary side  18   b  current also will flow through the inverse side of the MOSFET  34 , the current transformer  42  thus generating current in its secondary winding  42   b . This current brings about a voltage drop across the resistor  46  of a magnitude equal to the gate voltage of the MOSFET  34 . The value of the voltage drop is adjustable by the ratio between the two resistors  44 ,  46 . 
   MOSFET  34  is turned on during the time interval from t 1 , to t 2 , i.e. it is on also during the freewheeling phase. When current flows in the opposite direction the output voltage at the secondary side  42  of the current transformer  42  becomes negative and MOSFET  34  turns off. The second section L S2  of the power transformer  18  and the second MOSFET  36  (not shown in  FIG. 8 ) behave accordingly, yet with opposite sign. 
   Controlling synchronous rectifiers by means of current transformers, such as illustrated in  FIG. 8 , has certain disadvantages. On the one hand, a MOSFET requires a great current pulse to be turned on, which means that the windings ratio n2/n1 of the current transformer must be low. In the switched-on state, on the other hand, the gate current of the MOSFET is negligible, which means that a high windings ratio n2/n1 of the current transformer is required. 
   A description of prior art similar to what has been described above, but relating to a single phase forward transformer with synchronous rectification, including a current transformer, is to be found in “Synchronous Rectification Circuit Using A Current Transformer” by Y. Kubota et al., NTELC Conference Proceedings, September 2000, pages 267 to 273. 
   It is an object of the invention, starting from the state of the art as described above, to indicate a synchronous rectifier circuit for a push-pull voltage transformer that attains the fastest possible switching of the metal oxide semiconductor field effect transistors (MOSFETs) while, at the same time, causing the least possible power dissipation. This aim is to be reached, above all, by generating a higher switch-on current for the MOSFETs so as to keep the time of flow through the inverse diodes as short as possible, and of keeping the drive current as small as possible when the MOSFETs are in the on-state so as to minimize power dissipation. 
   SUMMARY OF THE INVENTION 
   The object is met by a synchronous rectifier circuit comprising the features recited in claim  1 . 
   In a synchronous rectifier circuit comprising a current converter of the kind described above, the invention more specifically provides for designing the transformer of the current converter such as to include first and second secondary windings to drive the MOSFET in two stages. The first secondary winding yields a relatively high current gain, while the second secondary winding yields a relatively low current gain. Therefore, the first secondary winding can be utilized for quickly turning on the MOSFET, the gate capacitance of the MOSFET being charged rapidly by the relatively great switch-on current. In the second stage, the MOSFET which has been turned on is kept in the on state with relatively low current gain, and that requires a smaller current. Thus the MOSFET is susceptible of being driven quickly and, when turned on, being kept with low loss in its switched state. 
   In the preferred embodiment of the invention, therefore, the electronic switches of the synchronous rectifier circuit are embodied by MOSFETs and the invention will be discussed below in the context of that type of transistors. However, the electronic switches likewise may be implemented as bipolar transistors or any other suitable type of switch. 
   In addition, the invention provides for a transconductance choke to reset the secondary side  42   b  of the current transformer  42 , the choke supporting the resetting or demagnetizing of the current transformer  42  after the switching operation. 
   While transconductance chokes are known in principle, they were not employed so far either in synchronous rectifiers or for resetting current transformers. The specific provision of the transconductance choke in the circuitry of the secondary side of the current transformer permits the resetting operation to be accomplished in clearly less time so that the freewheeling phase can be chosen to be shorter. That offers the advantage of permitting the push-pull voltage transformer, on the whole, to be operated at higher frequency and/or in the sense of greater variability of the clock ratio. As the freewheeling phase becomes shorter, the clock ratio of the transformer can be adjusted more flexibly, in response to the energy to be transmitted. A shorter freewheeling phase also contributes to increasing the efficiency. 
   Preferred embodiments of the invention are indicated in the dependent claims. 

   
     SHORT DESCRIPTION OF THE DRAWINGS 
     The invention will be described further by way of a preferred embodiment with reference to the drawings, in which: 
       FIG. 1  is a block diagram of a push-pull voltage transformer circuit in which the invention may be used; 
       FIG. 2  is a schematic circuit diagram of a push-pull voltage transformer, including a diode-type rectifier; 
       FIG. 3  shows idealized waveshapes of the output voltages and output currents of the push-pull voltage transformer shown in  FIG. 2 ; 
       FIG. 4  is an equivalent circuit diagram of a MOSFET which may replace the diodes shown in  FIG. 2 ; 
       FIG. 5  illustrates the replacement of the diode-type rectifier of  FIG. 1  by a synchronous rectifier including a MOSFET, in the form of a block diagram; 
       FIG. 6  is a schematic circuit diagram of the secondary side of a self-controlled synchronous rectifier; 
       FIG. 7  is a schematic circuit diagram of the secondary side of an IC controlled synchronous rectifier; 
       FIG. 8  is a schematic circuit diagram of the upper half of the secondary side of a synchronous rectifier including a current transformer, the lower half being mirror inverted; 
       FIG. 9  is a schematic circuit diagram of the upper half of the secondary side of a synchronous rectifier circuit according to the invention, the lower half being mirror inverted; 
       FIG. 10  is a schematic circuit diagram of the upper half of the secondary side of the synchronous rectifier circuit according to the invention, similar to the illustration in  FIG. 9  but with further details, in a first operating phase; 
       FIG. 11  shows a circuit diagram similar to  FIG. 10 , in a second operating phase; 
       FIG. 12  shows a circuit diagram similar to  FIGS. 10 and 11 , in a third operating phase; 
       FIG. 13  shows idealized waveshapes of the output voltages and output currents of the synchronous rectifier circuit according to  FIGS. 10 to 12 . 
   

   DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     FIG. 9  is a circuit diagram of the upper part of the secondary side of the synchronous rectifier according to the invention. The lower part of the secondary side is mirror inverted. The structure of the primary side may be of known design. 
   Components corresponding to those in the circuitry shown in  FIGS. 6 to 8  are designated by the same reference numerals. 
   At its secondary side, the synchronous rectifier circuit according to the invention comprises a first secondary winding section L s1  of the power transformer  18  connected in series with a current converter transformer  48  and a MOSFET. The current converter transformer  48  comprises a primary winding  48   a  and a secondary winding  48   b , the laffer being divided into two winding sections  48   b   1  and  48   b   2 . The first secondary winding section  48   b   1  is connected to a first branch which, in the embodiment illustrated in  FIG. 9 , includes a diode  50  and a transconductance choke  52 . The second secondary winding section  48   b   2  is connected to a second branch including a diode  54  and a resistor  56 . The transconductance choke  52  is also known as “magnetic amplifier” or “mag amp”. 
   The first and second branches are connected via another resistor  58  to a common reference potential of the synchronous rectifier circuit. 
   As is known and also shown in  FIG. 8 , a storage choke  28  and a storage capacitor  30  are provided to form an output filtering stage at the output of the synchronous rectifier circuit. 
   The function of the synchronous rectifier circuit, in principle, is based on the following switching behavior: To turn on the MOSFET  34 , a relatively high current is generated across the second secondary winding section  48   b   2 . At the same time, relatively low current flows through the first secondary winding section  48   b   1  and the first branch in which the transconductance choke  52  is located: .To that end, the winding ratio of the first and second winding sections  48   b   1 ,  48   b   2  to the primary winding  48   a  is suitably chosen. In particular, a low winding ratio n21/n1 in the order of magnitude of 10:1 of the second secondary winding section  48   b   2  to the primary winding  48   a  is chosen so that a high switch-on current will be generated and the MOSFET  34  be turned on very rapidly. A high winding ratio n22/n1 in the order of magnitude of 100:1 of the first secondary winding section  48   b   1  to the primary winding  48   a  together with the high inductivity of the transconductance choke then lead to slow current rise in the first branch. As the current increase, the inductivity of the transconductance choke  52  decreases—the transconductance choke  52  is “switched on”—and consequently the current across the resistor  56  becomes very small again. 
   This switching behavior is due to the property of the transconductance choke  52 , namely having a high inductivity at low current and acting like an open switch. As the current increases, the inductivity decreases until the transconductance choke acts like a closed switch. This switching behavior is exploited advantageously by the invention. 
   In practice, the switch-off behavior of the circuit is somewhat more complicated than in the other circumstances described above. In a practical implementation, therefore, some additional components are necessary and advantageous in combination with a non-ideal transconductance choke, as will be described below. 
     FIGS. 10 to 12  illustrate detailed circuit diagrams of the synchronous rectifier circuit according to the invention. The main constituent parts of the secondary rectifier continue to be the current transformer  48  and the MOSFET  34 , the current transformer  48  comprising a first output branch which includes the transconductance choke  52  and the diode  50 , and a second output branch which includes the diode  54  and the resistor  56 . In addition to the circuit of  FIG. 9 ,  FIGS. 10 to 12  show a diode  60  which is connected parallel to the resistor  56  and another diode  62  connected in series to the resistor  58 . The first and second branches connect the secondary winding  48   b  of the current transformer  48  through the diode  62  and resistor  58  and, furthermore, through a transistor  64  connected in parallel with them, especially a bipolar pnp-transistor, to the gate of the MOSFET  34 . 
     FIGS. 10 to 12  further show the transconductance choke  52  having a first winding  52   a  in the first branch and a second winding  52   b  connected in series to a diode  66  and a resistor  68 , this series connection being connected in parallel with the first and second branches, as may be seen in  FIGS. 10 to 12 . 
   Operation of the synchronous rectifier circuit according to  FIGS. 10 to 12  will be described in greater detail below, with reference to the waveshapes illustrated in  FIG. 13 . The idealized waveshapes of  FIG. 13  demonstrate the output voltage u 01  of the upper winding section L S1 , of the secondary side  18   b  of the power transformer  18 , furthermore the output current i 01 , of this winding section L S1 , and the drain-source voltage u DS  of the MOSFET  34 , as well as the reset current i RES  (see  FIG. 12 ) to reset the current converter transformer  48  and the transconductance choke  52 . 
   The output voltage u 01  is zero during the time interval from t 0  to t 1  (see  FIG. 13 ). During this phase, which corresponds to the freewheeling phase, both MOSFET switches  34 ,  36  are conducting (not shown in the figs but mirror inverted to the presentation in  FIGS. 10 to 12 ). 
   At time t 0  current i 01 , flows through the inverse diode of MOSFET  34  ( FIG. 10 ) and the current transformer  48 . Accordingly, current also flows through the diode  54  and begins to charge the input capacitance of the MOSFET  34 . This current depends in the first place on the winding ratio n21/n1 of the second secondary winding section  48   b   2  to the primary winding  48   a  of the current converter transformer  48  and on the load. 
   A negligibly small current also flows through the transconductance choke  52  during the first time interval TD (see  FIG. 13 ). The winding ratio n22/n1 of the second secondary winding section  48   b   2  to the primary winding  48   a  of the current converter transformer  48 , the transconductance choke  52  itself and the load all influence the switching behavior of the MOSFET  34  (see  FIG. 11 ). The winding ratio n22/n1 must be chosen depending on the transconductance choke  52  such that drive losses will be minimized. At the end of the time interval TD the transconductance choke  52  is saturated and the MOSFET  34  fully turned on (see  FIG. 13 ). 
   The diode  69  and the resistor  56  determine the level of the gate-source voltage of the MOSFET  34 . The resistor  58  serves to attenuate oscillations in the gate circuit. 
   At time t 1 , the MOSFET  34  continues to be conducting, while the other MOSFET  36  (not illustrated in the figs.) turns off. Consequently, all the output current flows through MOSFET  34 . 
   During the freewheeling phase, time interval t 2  to t 3 , the behavior of the circuit corresponds to what has been described above. Beginning at time t 3 , the voltage u 01  across the secondary winding section L S1 , becomes negative. Thereupon, the current through the MOSFET  34  decreases. As a consequence, the voltages across the secondary winding sections  48   b   1  and  48   b   2  of the current converter transformer  48  are inverted, the transistor  64  becomes conducting and, thereby, accelerates turning off of the MOSFET  34 . 
   The negative voltages at the secondary winding sections  48   b   1 ,  48   b   2  of the current transformer  48  start the resetting process of the transconductance choke  52 . A reset current i RES  flows through the resistor.  68  and the diode  66 , resetting the current transformer  48  and the transconductance choke  52  (see  FIG. 12 ). 
   The reset current is limited by the resistor  68 . The diode  66  makes sure that current will flow during this interval alone. Reset current i RES  stops to flow no later than at time  4  when the freewheeling phase begins. 
   If both MOSFETs  34 ,  36  should be non-conducting when the synchronous rectifier is switched on the respective inverse diodes take care of the flow of current when it begins to flow through the secondary winding  18   b  of the power transformer  18 . At the same time, current also flows through the primary winding  48   a  of the current converter transformer  48  and thus through the first and second secondary winding sections  48   b   1 ,  48   b   2 . The transconductance choke  52  prevents current from flowing through the first branch, but current serving to charge the gate capacitances of the MOSFETs  34 ,  36  does flow through the diode  54  of the second branch. The small transmission ratio in the order of 10:1 of the second secondary winding section  48   b   2  to the primary winding  48   a  of the current converter transformer  48  brings about a relatively great switching current and results in extreme acceleration of the driving of MOSFETs  34 ,  36 . This quick charging in turn cuts the time of current flow across the inverse diodes. The MOSFET  34 ,  36  becomes conducting as the threshold voltage is reached. In this context, the resistors  56 ,  58  serve as current limiters and the diode  60  as a voltage limiter. The transistor  24  is blocked at this time. As already described, also the transconductance choke  52  is blocked during this first switch-on phase. The transconductance choke  52  acts as a magnetic switch in this circuit. It is only after a certain charging time that the transconductance choke  52  becomes saturated, which corresponds to a closed switch condition, so that the gate capacitance of the MOSFET  34 ,  36  is fully charged through the diode  50  and thus the synchronous rectifier is fully controlled. 
   The selection of a greater transmission ratio in the order of 1:100 between the primary winding  48   a  and the first secondary winding section  48   b   1  of the current converter transformer  48  brings with it a smaller charging current and, therefore, less driving losses of the synchronous rectifier. 
   As already mentioned, it is an advantageous feature of the invention that the transconductance choke  52  and the current converter transformer  48  can be reset or demagnetized quickly and easily after a switching operation. To this end, the transconductance choke  52  is provided with a second winding  52   b , as may be seen in  FIGS. 10 to 12 , which may have four turns, for example. The winding direction, here, is opposed to that of the first winding  52   a  of the transconductance choke  52 . The second winding  52   b  is fed through both the resistor  68 , serving as current limiter, and the diode  66  preventing current to flow in the opposite direction. The reset operations of the transconductance choke  52  and the current converter transformer  48  each take place during the blocking phase of the MOSFETs  34 ,  36 . During this time period the voltage at the corresponding winding section of the secondary side  18   b  of the power transformer  18  is negative. Due to this negative potential, current will flow through the resistor  68 , diode  66 , transconductance choke  52  and secondary winding  48   b  of the current converter transformer  48 . The voltage at the secondary side of the current transformer jumps to negative potential, and the subsequent falling current flow magnetizes the current transformer and the transconductance choke in opposite direction. The return current i RES  breaks off (see  FIG. 13 ) as the voltage at the secondary side, e.g. L S1 , of the power transformer  18  returns to 0 volt (freewheeling phase). Thereafter, both inductivities, of the transconductance choke  51  as well as the current converter transformer  48 , are demagnetized and ready for the next switching operation. 
   The guidelines below may be followed for selecting and dimensioning the various components of the synchronous rectifier circuit according to the invention. 
   The power transformers, MOSFETs, in the first place are selected with a view to a low forward resistance, preferably a forward resistance R DS &lt;50 mΩ, a drain-source voltage and a drain-source current depending on the desired output voltage and output current. 
   The optimum transmission ratio of the current converter transformer should be determined experimentally, depending on the particular case of application. Winding ratios of the primary winding  48   a  to the first secondary winding section  48   b   1  to the second secondary winding section  48   b   2  of 1:50:5 and 1:100:10 have provided good test results in this configuration. 
   In the rectifier circuit, the transconductance choke  52  assumes the function of a switch. It is responsible for a second stage of driving the MOSFET to take place after a certain time TD, as described above. 
   A transconductance choke essentially consists of a saturable annular soft magnetic core including one or more windings. The hysteresis curve of the core is almost rectangular. The magnetic switch is either on or off, depending on the degree of magnetization of the choke. The behavior of the transconductance choke will be described briefly. When voltage is applied to the choke the inductivity of the choke, initially, is very high, and no current flows through the choke. After the lapse of a period of time TD, the choke enters into a state of saturation, and then its inductivity is very small. At this state, current flows through the winding of the transconductance choke, the magnetic switch is closed. The magnetic flux density remains almost constant as long as this current flows. When the flow of current through the choke is interrupted the magnetic field strength .decreases while the magnetic flux density continues to remain constant. If a current now flows in the opposite direction (reset current) or if a negative voltage is applied the inductivity of the choke goes down towards zero so that the transconductance choke will become demagnetized. If the reset current flows for a long enough time the transconductance choke will end up being fully demagnetized and thus be reset. In case the reset current continues to flow the transconductance choke will become magnetized in the opposite direction, and the magnetic switch is opened again. 
   The function of further individual components is as follows: 
   The diodes  50 ,  54 , and  58  prevent reset current from flowing during the blocking phase of the MOSFETs  34 ,  36 . Schottky diodes, for example, may be selected since their loss is rather small due to their small forward voltage and as they allow relative great currents to be applied. 
   The resistor  56  and the Zener diode  60  serve for adjusting the gate-source voltage. The resistor  58  serves as a gate resistor to suppress oscillations. Furthermore, the provision of the resistor  58  upstream of the transistor  64  abbreviates switching-off of-the MOSFET  34 , whereby the efficiency of the overall circuit is improved once more. The transistor  64  serves for discharging the gate capacitance of the MOSFETs  34 .  36 . The resistor  68  and the diode  66  serve for demagnetizing the transconductance choke  52  and the current transformer  48 . 
   The features disclosed in the specification above, in the claims and drawings may be significant for implementing the invention in its various embodiments, both individually and in any combination. 
   The invention provides a synchronous rectifier for use in a push-pull voltage transformer, being exceptionally efficient and quick in switching. This makes it possible, on the one hand, to apply the circuit according to the invention particularly also in multi-stage switch mode power supply and, at the same time, to obtain switch mode power supply operating at high frequency and/or being flexible as regards the clock ratio. The invention is particularly advantageous for switch mode power supply with low output voltage, such as &lt;24 V, where the losses of a diode-type rectifier are especially significant. Examples of applications are switch mode power supply for telecommunications systems, computers and industrial applications, as well as voltage supply for processor cores, and particularly all those applications which require low voltages and high currents. 
   LIST OF REFERENCE NUMERALS 
   
       
         10  input switching stage 
         12  power transformer 
         14  rectifier circuit 
         16  output filter 
         18  power transformer 
         18   a  primary side of the power transformer 
         18   b  secondary side of the power transformer 
         20  power transistor 
         22  power transistor 
         24  secondary diode 
         26  secondary diode 
         28  storage choke 
         30  storage capacitor 
         32  synchronous rectifier circuit 
         34  MOSFET, electronic switch 
         36  MOSFET, electronic switch 
         38  driver IC 
         40  driver IC 
         42  current transformer 
         42   a  primary winding of the current transformer 
         42   b  secondary winding of the current transformer 
         44  resistor 
         46  resistor 
         48  current converter transformer 
         48   a  primary winding of current converter transformer 
         48   b  secondary winding of current converter transformer 
         48   b   1  first secondary winding section 
         48   b   2  second secondary winding section 
         50  diode 
         52  transconductance choke 
         52   a  first winding of the transconductance choke 
         52   b  second winding of the transconductance choke 
         54  diode 
         56  resistor 
         58  resistor 
         60  diode 
         62  diode 
         64  transistor 
         66  diode 
         68  resistor