Abstract:
A direct power amplifier modulator that accepts bipolar I and Q data, either baseband or multi-carrier digital IF, and generates the resultant magnitude and sign components. Also described is a direct power amplifier modulator that generates quadrature terms of the carrier frequency offset by 90°, wherein each term is shifted by 180° under control of the respective sign components of the quadrature I and Q samples. The present invention also provides a transistor structure configured to accept differential quadrature frequency components that are combined in a post PA outphasing combiner.

Description:
CROSS-REFERENCE TO RELATED APPLICATION AND CLAIM OF PRIORITY 
   The present invention is related to that disclosed in U.S. Provisional Patent No. 60/616,912, filed Oct. 7, 2004, entitled “Apparatus And Method For Direct Quadrature Power Amplifier Modulation”. U.S. Provisional Patent No. 60/616,912 is assigned to the assignee of the present application. The subject matter disclosed in U.S. Provisional Patent No. 60/616,912 is hereby incorporated by reference into the present disclosure as if fully set forth herein. The present application hereby claims priority under 35 U.S.C. §119(e) to U.S. Provisional Patent No. 60/616,912. 

   TECHNICAL FIELD OF THE INVENTION 
   The present invention relates generally to RF transmitters and, in particular, to a switch mode RF power amplifier. 
   BACKGROUND OF THE INVENTION 
   In conventional radio frequency (RF) transmitters, RF power amplifiers are operated linearly in class A, class AB, or class C bias conditions. The problem with these bias classes is that the efficiency of the power stage is limited due to high dissipation in the power stage resulting from either quiescent bias or basic ohmic losses. A large body of research has been devoted to the development of new RF transmitters capable of generating linear RF power using non-linear power amplifiers operating in switch mode in order to increase the power-added efficiency. 
   Various switch mode techniques have been proposed to improve the power-added efficiency of the power amplifier (PA) stage. These techniques include: i) polar modulation; ii) envelope elimination and restoration; iii) LINC; and iv) delta sigma modulation. In each case, some form of direct modulation is applied in the power amplifier stage. The general concept is to operate the final amplifier stage in switch mode, wherein the final power transistor stage is driven between pinch-off and saturation at the carrier frequency rate or some multiple thereof. By minimizing the percentage of a cycle in which the transistor is in the linear operating region, the power dissipated in the transistor is minimized and a high level of power added efficiency is obtained. 
   The obvious drawback of this approach is that, to maintain high modulation accuracy and to avoid generating adjacent channel interference, a high degree of envelope linearity is required in the amplifier stage. As a result, high efficiency power amplification techniques focus on maintaining the envelope integrity of the amplified waveform, although the carrier itself will be clipped by switch mode operation. The result is that the in-band modulation accuracy and adjacent channel integrity are preserved, even though a high level of harmonics are generated. The harmonic content is easily filtered after the amplifier stage. 
   In the polar modulation (or EER) method, the quadrature baseband information is converted to polar components of amplitude and phase. An oscillator operating at the carrier frequency drives the gate of a switch-mode power amplifier (PA) with a constant amplitude, constant frequency signal. The phase component of the complex waveform is used to phase modulate the oscillator controlling the phase of the PA output. The amplitude component is used to amplitude modulate the switch-mode PA by controlling the PA drain bias. Pre-distortion is commonly used to maintain a linear envelope. A high level of carrier harmonics are generated by the switching waveform, so a greater amount of harmonic filtering must be used than is typically required after a linear PA. 
   One weakness of the polar modulation approach is that the delays through the amplitude modulation path and the phase modulation path are inherently different due to fundamentally different circuit topologies and are subject to differing amounts of variation due to temperature or component variations. A small timing error (&gt;2 nanoseconds) may result in intolerable modulation errors for wideband modulation formats. 
   In the LINC method, two common frequency components generated from switch mode power amplifier stages are phase modulated then combined so they add and subtract linearly to produce a quadrature modulated carrier. This requires some unique power combining techniques that enable combination of non-coherent waveforms into a load without excessive power dissipation in the combiner. Achieving high efficiency combining has been a significant challenge to implementation of this technique. 
   The delta-sigma modulation method typically uses a high-order (fourth) delta sigma bandpass loop as a single bit analog to digital converter. This produces a noise transfer function having a notch at the Fs/2 or Fs/4 frequency. A good signal-to-noise ratio may therefore be achieved when a carrier is operated within the notch of the noise transfer function. The single bit output of this loop is used to drive the gate of a switch mode PA. A narrow bandpass filter is used after the power amplifier (PA) to eliminate the broadband noise outside the carrier bandwidth. 
   One of the challenges of implementing this approach is that the delta-sigma loop must be clocked at twice to four times the carrier frequency. For carriers in the US PCS or IMT-2000 bands, this can be a significant challenge, given current device speeds. Another limitation is that the instantaneous bandwidth of the modulator is limited to about 20 MHz, depending on the order and clock rate of the delta-sigma loop. In addition, the fixed bandpass filter following the PA stage greatly limits the operating band of the PA. 
   Therefore, there is a need in the art for an improved power amplifier that performs switch mode power amplification without encountering the problems described above. 
   SUMMARY OF THE INVENTION 
   The present invention provides a form of switch mode power amplification that avoids the problems encountered with the prior methods. The present invention provides a direct power amplifier modulator that accepts bipolar I and Q data, either baseband or multi-carrier digital IF, and generates the resultant magnitude and sign components. The present invention also provides a direct power amplifier modulator that generates quadrature terms of the carrier frequency offset by 90°, wherein each term is shifted by 180° under control of the respective sign components of the quadrature I and Q samples. The present invention also provides a transistor structure configured to accept differential quadrature frequency components that are combined in a post PA outphasing combiner. 
   According to one embodiment of the present invention, there is provided an RF transmitter comprising a quadrature source data block capable of receiving an input in-phase (I) data stream and an input quadrature (Q) data stream and generating therefrom an in-phase absolute value (|I|′) data signal and a quadrature absolute value (|Q|′) data signal; and a quadrature modulation block comprising a first totem pole transistor structure containing a first bias modulator controlled by said |I|′ data signal and a second totem pole transistor structure containing a second bias modulator controlled by said |Q|′ data signal. 
   According to other embodiments of the present invention, there are provided base and mobile stations for use in a wireless network, each for use in a wireless network, each comprising a processor and an accessible memory; a transceiver connected to be controlled by the processor, the transceiver including a quadrature source data block capable of receiving an input in-phase (I) data stream and an input quadrature (Q) data stream and generating therefrom an in-phase absolute value (|I|′) data signal and a quadrature absolute value (|Q|′) data signal, and a quadrature modulation block comprising a first totem pole transistor structure containing a first bias modulator controlled by said |I|′ data signal and a second totem pole transistor structure containing a second bias modulator controlled by said |Q|′ data signal. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention and its advantages, reference is now made to the following description taken in conjunction with the accompanying drawings, in which like reference numerals represent like parts: 
       FIG. 1  illustrates exemplary wireless network according to the principles of the present invention; 
       FIG. 2  illustrates exemplary base station according to an exemplary embodiment of the present invention; 
       FIG. 3  illustrates wireless mobile station according to an advantageous embodiment of the present invention; 
       FIG. 4  illustrates a quadrature data converter block according to the principles of the present invention; and 
       FIG. 5  illustrates a quadrature drive of an H-bridge power amplifier according to an exemplary embodiment of the present invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIGS. 1 through 5 , discussed below, and the various embodiments used to describe the principles of the present invention in this patent document are by way of illustration only and should not be construed in any way to limit the scope of the invention. Those skilled in the art will understand that the principles of the present invention may be implemented in any suitably arranged RF transmitter. 
   The invention disclosed by this form attempts to gain the efficiency advantages of the other switch modes of power amplifier operation while avoiding the issues associated with the these alternate approaches. The technique described herein involves a direct quadrature modulation of the switch-mode power amplifier. 
     FIG. 1  illustrates exemplary wireless network according to the principles of the present invention. 
   Wireless network  100  comprises a plurality of cell sites  121 - 123 , each containing one of the base stations, BS  101 , BS  102 , or BS  103 . Base stations  101 - 103  communicate with a plurality of mobile stations (MS)  111 - 114  over code division multiple access (CDMA) channels according to, for example, the IS-2000 standard (i.e., CDMA2000). In an advantageous embodiment of the present invention, mobile stations  111 - 114  are capable of receiving data traffic and/or voice traffic on two or more CDMA channels simultaneously. Mobile stations  111 - 114  may be any suitable wireless devices (e.g., conventional cell phones, PCS handsets, personal digital assistant (PDA) handsets, portable computers, telemetry devices) that are capable of communicating with base stations  101 - 103  via wireless links. 
   The present invention is not limited to mobile devices. The present invention also encompasses other types of wireless access terminals, including fixed wireless terminals. For the sake of simplicity, only mobile stations are shown and discussed hereafter. However, it should be understood that the use of the term “mobile station” in the claims and in the description below is intended to encompass both truly mobile devices (e.g., cell phones, wireless laptops) and stationary wireless terminals (e.g., a machine monitor with wireless capability). 
   Dotted lines show the approximate boundaries of cell sites  121 - 123  in which base stations  101 - 103  are located. The cell sites are shown approximately circular for the purposes of illustration and explanation only. It should be clearly understood that the cell sites may have other irregular shapes, depending on the cell configuration selected and natural and man-made obstructions. 
   As is well known in the art, each of cell sites  121 - 123  is comprised of a plurality of sectors, where a directional antenna coupled to the base station illuminates each sector. The embodiment of  FIG. 1  illustrates the base station in the center of the cell. Alternate embodiments may position the directional antennas in corners of the sectors. The system of the present invention is not limited to any particular cell site configuration. 
   In one embodiment of the present invention, each of BS  101 , BS  102  and BS  103  comprises a base station controller (BSC) and one or more base transceiver subsystem(s) (BTS). Base station controllers and base transceiver subsystems are well known to those skilled in the art. A base station controller is a device that manages wireless communications resources, including the base transceiver subsystems, for specified cells within a wireless communications network. A base transceiver subsystem comprises the RF transceivers, antennas, and other electrical equipment located in each cell site. This equipment may include air conditioning units, heating units, electrical supplies, telephone line interfaces and RF transmitters and RF receivers. For the purpose of simplicity and clarity in explaining the operation of the present invention, the base transceiver subsystems in each of cells  121 ,  122  and  123  and the base station controller associated with each base transceiver subsystem are collectively represented by BS  101 , BS  102  and BS  103 , respectively. 
   BS  101 , BS  102  and BS  103  transfer voice and data signals between each other and the public switched telephone network (PSTN) (not shown) via communication line  131  and mobile switching center (MSC)  140 . BS  101 , BS  102  and BS  103  also transfer data signals, such as packet data, with the Internet (not shown) via communication line  131  and packet data server node (PDSN)  150 . Packet control function (PCF) unit  190  controls the flow of data packets between base stations  101 - 103  and PDSN  150 . PCF unit  190  may be implemented as part of PDSN  150 , as part of MSC  140 , or as a stand-alone device that communicates with PDSN  150 , as shown in  FIG. 1 . Line  131  also provides the connection path for control signals transmitted between MSC  140  and BS  101 , BS  102  and BS  103  that establish connections for voice and data circuits between MSC  140  and BS  101 , BS  102  and BS  103 . 
   Communication line  131  may be any suitable connection means, including a T 1  line, a T 3  line, a fiber optic link, a network packet data backbone connection, or any other type of data connection. Line  131  links each vocoder in the BSC with switch elements in MSC  140 . The connections on line  131  may transmit analog voice signals or digital voice signals in pulse code modulated (PCM) format, Internet Protocol (IP) format, asynchronous transfer mode (ATM) format, or the like. 
   MSC  140  is a switching device that provides services and coordination between the subscribers in a wireless network and external networks, such as the PSTN or Internet. MSC  140  is well known to those skilled in the art. In some embodiments of the present invention, communications line  131  may be several different data links where each data link couples one of BS  101 , BS  102 , or BS  103  to MSC  140 . 
   In the exemplary wireless network  100 , MS  111  is located in cell site  121  and is in communication with BS  101 . MS  113  is located in cell site  122  and is in communication with BS  102 . MS  114  is located in cell site  123  and is in communication with BS  103 . MS  112  is also located close to the edge of cell site  123  and is moving in the direction of cell site  123 , as indicated by the direction arrow proximate MS  112 . At some point, as MS  112  moves into cell site  123  and out of cell site  121 , a hand-off will occur. 
     FIG. 2  illustrates exemplary base station  101  in greater detail according to an exemplary embodiment of the present invention. Base station  101  comprises base station controller (BSC)  210  and base transceiver station (BTS)  220 . Base station controllers and base transceiver stations were described previously in connection with  FIG. 1 . BSC  210  manages the resources in cell site  121 , including BTS  220 . BTS  120  comprises BTS controller  225 , channel controller  235  (which contains representative channel element  240 ), transceiver interface (IF)  245 , RF transceiver  250 , and antenna array  255 . 
   BTS controller  225  comprises processing circuitry and memory capable of executing an operating program that controls the overall operation of BTS  220  and communicates with BSC  210 . Under normal conditions, BTS controller  225  directs the operation of channel controller  235 , which contains a number of channel elements, including channel element  240 , that perform bi-directional communications in the forward channel and the reverse channel. A “forward” channel refers to outbound signals from the base station to the mobile station and a “reverse” channel refers to inbound signals from the mobile station to the base station. Transceiver IF  245  transfers the bi-directional channel signals between channel controller  240  and RF transceiver  250 . 
   Antenna array  255  transmits forward channel signals received from RF transceiver  250  to mobile stations in the coverage area of BS  101 . Antenna array  255  also sends to RF transceiver  250  reverse channel signals received from mobile stations in the coverage area of BS  101 . In a preferred embodiment of the present invention, antenna array  255  is multi-sector antenna, such as a three-sector antenna in which each antenna sector is responsible for transmitting and receiving in a 120° arc of coverage area. Additionally, RF transceiver  250  may contain an antenna selection unit to select among different antennas in antenna array  255  during both transmit and receive operations. 
     FIG. 3  illustrates wireless mobile station  111  according to an advantageous embodiment of the present invention. Wireless mobile station  111  comprises antenna  305 , radio frequency (RF) transceiver  310 , transmit (TX) processing circuitry  315 , microphone  320 , and receive (RX) processing circuitry  325 . MS  111  also comprises speaker  330 , main processor  340 , input/output (I/O) interface (IF)  345 , keypad  350 , display  355 , and memory  360 . Memory  360  further comprises basic operating system (OS) program  361 . 
   Radio frequency (RF) transceiver  310  receives from antenna  305  an incoming RF signal transmitted by a base station of wireless network  100 . Radio frequency (RF) transceiver  310  down-converts the incoming RF signal to produce an intermediate frequency (IF) or a baseband signal. The IF or baseband signal is sent to receiver (RX) processing circuitry  325  that produces a processed baseband signal by filtering, decoding, and/or digitizing the baseband or IF signal. Receiver (RX) processing circuitry  325  transmits the processed baseband signal to speaker  330  (i.e., voice data) or to main processor  340  for further processing (e.g., web browsing). 
   Transmitter (TX) processing circuitry  315  receives analog or digital voice data from microphone  320  or other outgoing baseband data (e.g., web data, e-mail, interactive video game data) from main processor  340 . Transmitter (TX) processing circuitry  315  encodes, multiplexes, and/or digitizes the outgoing baseband data to produce a processed baseband or IF signal. Radio frequency (RF) transceiver  310  receives the outgoing processed baseband or IF signal from transmitter (TX) processing circuitry  315 . Radio frequency (RF) transceiver  310  up-converts the baseband or IF signal to a radio frequency (RF) signal that is transmitted via antenna  305 . 
   In an advantageous embodiment of the present invention, main processor  340  is a microprocessor or microcontroller. Memory  360  is coupled to main processor  340 . According to an advantageous embodiment of the present invention, part of memory  360  comprises a random access memory (RAM) and another part of memory  360  comprises a Flash memory, which acts as a read-only memory (ROM). 
   Main processor  340  executes basic operating system (OS) program  361  stored in memory  360  in order to control the overall operation of wireless mobile station  111 . In one such operation, main processor  340  controls the reception of forward channel signals and the transmission of reverse channel signals by radio frequency (RF) transceiver  310 , receiver (RX) processing circuitry  325 , and transmitter (TX) processing circuitry  315 , in accordance with well-known principles. 
   Main processor  340  is capable of executing other processes and programs resident in memory  360 . Main processor  340  can move data into or out of memory  360 , as required by an executing process. Main processor  340  is also coupled to I/O interface  345 . I/O interface  345  provides mobile station  111  with the ability to connect to other devices such as laptop computers and handheld computers. I/O interface  345  is the communication path between these accessories and main controller  340 . 
   Main processor  340  is also coupled to keypad  350  and display unit  355 . The operator of mobile station  111  uses keypad  350  to enter data into mobile station  111 . Display  355  may be a liquid crystal display capable of rendering text and/or at least limited graphics from web sites. Alternate embodiments may use other types of displays. 
     FIG. 4  illustrates a quadrature data converter block according to the principles of the present invention. In the direct quadrature modulation method disclosed herein, the switch-mode PA consists of two transistor structures—one for the I channel amplification and one for the Q channel. The two paths are combined into the load circuit to form the complete vector. In this method, the interpolated digital I and Q baseband information is conditioned in data format converters  405 / 430  to create the |I| and |Q| signals which represent the absolute value of the bipolar I and Q signals. The sign information of the I and Q data is separated into the sgn(I) and sgn(Q) signals. These signals are essentially the sign bits of the baseband I and Q data. 
   Similar to the polar modulation approach, an oscillator  425  generates the basic carrier frequency, f c . This is generated in digital form in both the 0° component and a 90° delayed version. The complementary terms of the 0° component are generated from an exclusive-OR (XOR) function  420  that is also driven by the sgn(I) bit. This bit causes an inversion in the XOR function  420  whenever the baseband I data is a negative value. Likewise, the complementary terms of the 90° component are generated from an This causes an inversion in the XOR function  435  whenever the baseband Q data is a negative value. 
   In this manner the negative I and Q values can be modulated from a unipolar PA circuit. The complementary f c (0°) signals drive a totem-pole transistor structure to form the I switch. The complementary f c (90°) signals drive a totem-pole transistor structure to form the Q switch. Therefore, there is no actual phase information that must be modulated onto the carrier. The phases of the I and Q waveforms are constant with the exception of switching between the 0° and 180° states when the original I and Q waveform transition from positive to negative values. 
   |I| is passed through linearizer  410  and to DAC  415 , to produce |I|′. |I|′ is passed to low pass filter  417 . Similarily, |Q| is passed through linearizer  440  and to DAC  445 , to produce |Q|′. |Q|′ is passed to low pass filter  447 . 
     FIG. 5  illustrates a quadrature drive of an H-bridge power amplifier according to an exemplary embodiment of the present invention. The |I|′ and |Q|′ signals drive an amplitude/bias modulator block  505 / 555 . This block is in essence a DC-to-DC converter that accepts a constant bias voltage and generates DC current at a secondary voltage that is proportional to the |I|′ and |Q|′ control signals. 
   The variable output voltages of the amplitude modulator blocks supply the drain bias of the power amplifiers stages ( 510 / 515 / 520 / 525 / 530 / 535 / 540  and  560 / 565 / 570 / 575 / 580 / 585 / 550 , respectively). This results in a unipolar amplitude modulation of both the I and Q branches. Finally the phase and amplitude components of the modulation vector are then combined at the bandpass filter  590  to form the complete vector across RL  595 . 
   Obviously, a perfectly linear amplitude and phase response will not be obtain from the direct drain modulation of the final transistor stages. Consequently, a linearization function is used to pre-distort the I and Q waveforms to compensate for the non-linear transistor response so that a linear amplitude modulation is obtained. The I and Q waveforms are pre-distorted so that after combination the composite vector has high modulation accuracy and low adjacent channel interference. 
   Due to the switch mode operation, high harmonic levels will be generated by the PA stage. The post-amplifier bandpass filter (BPF) attenuates the high harmonic levels resulting from the switching transients. In this manner, the switch-mode PA is able to deliver highly efficient operation using a quadrature modulation approach that is robust in its delay differences. 
   One advantage of the quadrature modulation technique is that since the I and Q conditioning circuits can be implemented identically on a common semiconductor, the delay differences would be negligible. The main timing constraint in the direct quadrature technique is that the sgn(I) be held synchronous with the |I| signal and likewise the sgn(Q) be held synchronous with the |Q| signal. This would be a simple matter since these functions would be implemented digitally on a common silicon substrate. This avoids the phase and magnitude problems commonly encountered in polar modulation techniques which tend to limit the practical bandwith of that technique. 
   Another advantage is that, since the interface to most wireless modems is typically a digital quadrature interface, there is no quadrature-to-polar converter block that must be implemented in this architecture. The quadrature modulator generates the magnitude and sign components of the I and Q terms based solely on I and Q amplitude without having to calculate phase. This is an advantage over the polar modulation and LINC techniques which both rely on phase modulation of the carrier. 
   The direct quadrature modulation technique has advantages over the delta-sigma modulation approach in that there are no issues with loop stability. The fundamental clock used to derive the carrier frequency term is also easier to generate with the direct quadrature modulation technique, since it is generated at the fundamental carrier frequency rather than at two or four times the frequency. For carriers within the 800 to 2200 MHz range, this makes the direct quadrature modulation technique much more feasible for implementation in high-speed digital logic. Current semiconductor processes such as CMOS will support these speeds, whereas clock rates at 8 GHz will require more expensive device technologies such as SiGe BiCMOS. 
   The direct quadrature modulation technique also has advantages over the delta-sigma modulation approach in that it does not contain the severe bandwidth limitations caused by the noise transfer function of high-order delta sigma loops. Also, the bandpass filter used in the direct quadrature modulation technique can be a wideband filter covering the entire operating band of interest, whereas the delta-sigma modulator approach typically must utilize a narrow band filter that is constrained by the width of the delta-sigma modulator noise transfer function. This bandwidth will typically be on the order of 20 MHz or less, which is about a 1 to 2% bandwidth. This is a severe limitation for wireless operators that own different frequency blocks in various markets that want to avoid stocking multiple versions of the same power amplifier. This would not be an issue for a power amplifier using the direct quadrature modulation technique. 
   Although the present invention has been described with an exemplary embodiment, various changes and modifications may be suggested to one skilled in the art. It is intended that the present invention encompass such changes and modifications as fall within the scope of the appended claims.