Abstract:
Adaptive rail power amplifier technology processes an audio signal by feeding the audio signal to the power amplifier to produce an output signal, applying positive and negative power supply voltages centered with respect to the audio signal to the positive and negative power supply rails of the power amplifier, comparing the output signal with the positive and negative power supply rail voltages to produce dynamically varying positive and negative control signals, feeding the positive and negative control signals to positive and negative high current charge pumps and adding supplemental positive and negative voltages from the positive and negative charge pumps to the positive and negative power supply rails to produce a linear adaptive rail voltage which tracks the output signal.

Description:
CROSS-REFERENCE TO PENDING APPLICATIONS 
     This application claims priority to U.S. Provisional Patent Application No. 61/622,696, filed on Apr. 11, 2012. 
    
    
     BACKGROUND OF THE INVENTION 
     Numerous designs have been developed and used over the past years incorporating various methods of providing a modulated power supply for an audio amplifier in order to improve efficiency and reduce dissipation of the power devices. Many prior art designs have disclosed either dual or multiple rails that switch to a higher voltage when the output swing of the amplifier is near clipping. These designs have been termed Class H and Class G where a secondary or multiple rail voltages are selected as required based on amplifier output swing. More complex designs have been realized with continuously variable rails that track the input signal and adjust the power supply rails so as to maintain a constant voltage between the output devices output swing and the power supply rail voltage. Most of these amplifiers are very complex requiring Pulse Width Modulation of the power supply and increased manufacturing requirements due to the large associated circuitry required to provide the tracking supply rails. Countless Class D designs have also been offered commercially which convert an input audio signal into a series of output pulses. When the pulses are averaged over time and low pass filtered to remove higher order harmonic information the output will be a replica of the input signal. While Class D offers the highest level of efficiency it is also one of the most difficult to use in applications where low EMI/RFI performance is required. While all of the various topologies have seen varying degrees of success commercially, the designs that offer the best cost vs. performance gain the widest market acceptance. Many of the numerous designs have excellent performance but may also be the most difficult to manufacture. At the same time high output automotive audio power amplifiers based on switch mode power supply technology has been available for years as aftermarket products but have not been embraced by the original equipment manufacturers (OEM) due to a number of undesirable side effects including switching transients which cause large levels of RFI emissions. In order to deliver high power automotive audio systems an efficient, high power DC to DC converter is required, which will convert the 12 volt automotive battery voltage to a higher supply voltage with high current output capability. The automotive charging system typically produces between 13.5 and 14.4 volts when the engine is running. While this is a slight increase above the 12 volt battery voltage this is not enough for high power amplification of audio signals. Amplifiers capable of high output power either need more voltage swing than the typical 14.4 volts available with the engine running or need to provide an extremely high current output in order to drive very low impedance loads. Typically, speakers with lower impedances have lower efficiencies and therefore a gain in output power with audio amplifiers that can deliver higher output current may not result in a large net gain in sound pressure levels. The formula to calculate output power is given by: 
     
       
         
           
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     An ideal power amplifier that can swing all the way to the power supply rails with a 14.4 volt supply can deliver a peak amplitude of 7.2 volts when connected to a 4 ohm load and would deliver 6.48 watts. Most automotive audio power amplifiers are dual amplifiers connected in what is termed “Bridge Mode” with one amplifier channel swinging positive and one swinging negative with the load connected between the two amplifier outputs and can, as a result, deliver twice the voltage swing across the load. This means that an ideal amplifier that can swing to the rails in bridge mode can deliver 14.4 volts peak which would deliver a total of 25.9 watts into a 4 ohm load. In reality, most power amplifiers are far from ideal and can typically only swing to within about 1.5 volts of the positive and negative power supply rails. As a result, the real world power amplifiers actual power output with the alternator running and 14.4 volts at the battery is closer to 16 watts. Thus it becomes obvious that the output of a car audio amplifier is limited by the voltage of the car battery with the alternator running. In most actual car systems, the amplifiers are connected in bridge mode configuration as described above, and speaker impedances are no higher than 4Ω, but it becomes apparent that the maximum output power per channel is roughly 30 watts even when driving a 2 ohm load and only about 16 watts with a 4 ohm load. High-power car amplifiers have been available for many years in the automotive aftermarket and these amplifiers use a DC-to-DC converter to generate a higher power supply voltage. In order to increase the battery voltage to a level capable of producing a higher power level most aftermarket automotive power amplifiers use switch mode power supplies SMPS to convert or transfer power from the 12 volt automotive battery (14.4 volts with the engine running) to a higher output voltage. While switching power supplies have seen improvements in terms of output power and efficiency even the best designs today produce an unacceptable level of radio frequency interference RFI and as a result have not seen wide acceptance for use in OEM vehicles. Other improvements in SMPS have been made offering higher switching frequencies, which allow component sizes to be reduced but produce even higher levels of RFI emissions. The common (SMPS) used in automotive aftermarket audio applications switches the battery voltage at a frequency between 25 kHz and 100 kHz to generate an AC square wave signal at the primary side of a step-up transformer. The stepped up waveform on the secondary of the transformer is rectified and filtered back to a DC signal. The output is typically a symmetrical +/−25 to 35 volts. 
     DC-DC converters based on charge pump or flying capacitor technology have been widely used in low power DC-DC converters but have seen limited use in high power applications due to a number of limitations including high pulse currents that occur at the switching transients which reduce efficiency and increase RFI problems. The low power circuits typically switch at higher frequencies between 20 KHz and 150 KHz which reduce the size of the capacitors but also contribute to an increase the RFI emissions. Integrated circuits have been produced for years based on the concepts of charge pump circuitry and have provided circuits which offer low current designs capable of delivering only milliamps of current to an external load. U.S. Pat. No. 5,066,871 is an example of one such design but many of the integrated circuit manufacturers offer IC&#39;s based on charge pump technology. There are many current offerings for low power switched capacitor technology in integrated circuit form from manufacturers including Analog Devices, Linear Technologies and National Semiconductor, to name just a few. However, none of these circuits can be used in a higher power application that can deliver amperes of output power required for automotive audio power amplification. 
     One recent prior art system offers improved switched capacitor technology by charging a capacitor to the supply voltage and switching the charged capacitor when additional output swing is required. In order to keep the amplifier output swing centered, a reference voltage is added to the input to switch the amplifier center bias when the additional capacitor voltage is switched on. One drawback to this system is the adding of undesirable switching transients in the output signal. While this system will double the power supply voltage when needed, it is also limited to two times the power supply voltage, i.e. 28 volts with a 14 volt supply, and therefore requires relatively low impedance drivers in order to gain large amounts of output power. This system operates as a class H or class G amplifier when the additional rail voltage is switched on and off, which improves dissipated output device heat but does not gain the full advantage of a tracking rail or adaptive rail design. A full tracking rail or adaptive rail design will provide even better reduction of dissipation. The implementation of a pulsed rail configuration will greatly reduce the heat dissipation of the power MOSFETs used in switching the supply voltage. 
     In another recent prior art system, a power amplifier is centered between the supply voltage and both the positive and negative rails are increased as the output of the amplifier requires more voltage swing. This system basically uses additional power amplifiers with a gain of 1 to track the audio power amplifier output at unity gain. The system monitors the audio amplifiers input or output swing and when a threshold is exceeded the power boost amplifiers will boost the power supply rails by driving charged capacitors between the output of the boost amplifiers and the power supply rails. This system does provide a tracking rail design which tracks the output swing of the audio power amplifier. However, the net gain in output power is relatively small, on the order of a few watts, compared to bridge mode designs. This is so because it is a single ended design. The second major drawback to this design is the complexity of the system, including the power boost amplifiers. In order to provide boost amplifiers with unity gain, a full amplifier circuit is implemented with a power MOSFET output stage. A relatively small gain in total system output power is achieved at the expense of increased cost and complexity in implementing this system. While the power boost amplifiers certainly can be fully integrated in IC form, reducing build complexity, the design requires large output MOSFETs and will therefore require expensive integrated circuit packaging that will provide some form of heat sink interface to keep the output devices cool. The prior art system also teaches using a gain slightly above unity in order to avoid capacitor voltage droop or sag due to the capacitor discharging under amplifier load. The amount of voltage droop or sag is difficult to measure in operation and, therefore, adding additional gain may be inadequate in some circumstances or may be excessive in others. 
     It is, therefore, an object of the invention to provide an adaptive rail power amplifier using charge pump circuitry which overcomes the limitations of the above-mentioned prior art designs allowing use in OEM automotive vehicles. It is another object to provide an adaptive rail power amplifier with the ability to produce an output voltage swing above the input power supply voltage dynamically and as needed. In particular, it is an object of the invention to provide a power amplifier with adaptive tracking power supply rails offering greatly reduced complexity. It is a further object of the invention to provide an adaptive rail power amplifier with dynamically controlled charge pump converters for use in automotive power amplifiers capable of delivering well in excess of 100 watts of output power. It is a further object of the invention to provide an increase of up to 3 times the input supply voltage as required, to increase the amplifier output swing. It is a yet a further object of the invention to provide an adaptive rail power amplifier technology capable of high power output level for use in OEM automotive applications without causing problems in radio reception. It is still another objective of the invention to provide an adaptive rail power amplifier technology with higher efficiency and reduced heat dissipation. It is yet another objective of the invention to offer improved tracking of the power supply rails that can automatically adapt to any droop or sag voltage in the stored charge of the capacitor used to elevate the power supply rails. 
     It is also an object of the invention to provide an alternate embodiment of the adaptive rail power amplifier which has the same level of performance as the above described invention with reduced parts and complexity. It is a further object of the invention to provide an alternate embodiment with a simplified design which lends itself to full integration. It is another object of the alternate embodiment of the invention to require a single reference voltage for proper operation and signal tracking. 
     SUMMARY OF THE INVENTION 
     With the forgoing and other objects in view there is provided, in accordance with the invention, an adaptive rail power amplifier technology that compares the output signal of the power amplifier with positive and negative adaptive power supply rails generating a pulsed control signal fed to high current charge pump circuits for providing an output voltage that is capable of exceeding the input supply voltage when the output signal swings beyond the limits of the input voltage. 
     The adaptive rail high current charge pump circuit includes a charge storage device, also called a flying capacitor, with one of its plates connected to two high current switches. One high current switch is connected to the supply through a ferrite bead. The other high current switch is connected to ground through another ferrite bead. The other plate of the flying capacitor is connected to two high current diodes. The first high current diode is connected to the battery. The other high current diode is connected to the output node and to second charge storage capacitor to smooth the output voltage and deliver output current to the load. A dead zone circuit generates the proper timing to control the high current switches. The dead zone outputs are connected to the gate control of the high current switches. The charge pump circuit is controlled by the output of a comparator which compares the output signal of the power amplifier with the rail voltage, producing a dynamically varying pulsed output signal in response to the output swing of the amplifier and the adaptive rail output of the charge pump circuit. 
     In another embodiment of the invention, an adaptive rail control signal is derived by taking the difference between the amplifier output signal and the rail voltage. The derived difference signal is used to control a charge pump circuit in both saturated and linear mode, producing a linear adaptive rail voltage which tracks the output signal so as to provide an adaptively increasing power supply rail voltage. 
     In accordance with an added feature of the invention, multiple sections or blocks of the charge pump circuitry are combined to provide even higher output voltages, thereby increasing further the output voltage swing of the amplifier and allowing the use of higher efficiency higher impedance speaker loads. 
     While the invention will be described for use in automotive sound systems it is understood that many other applications for the invention are possible including any application where limited power supply voltage is seen. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Other objects and advantages of the invention will become apparent upon reading the following detailed description and upon reference to the drawings in which: 
         FIG. 1  is a block diagram of a typical prior art car audio system; 
         FIG. 2  is a simplified schematic of the typical switch mode power supply used in prior art aftermarket audio systems; 
         FIG. 3  is a block diagram of a pulsed mode adaptive rail power amplifier embodiment of the invention for use in automotive applications; 
         FIG. 4  is a partial block, partial schematic diagram of the positive charge pump circuit of  FIG. 3 ; 
         FIG. 5  is a partial block, partial schematic diagram of the negative charge pump circuit of  FIG. 3 ; 
         FIG. 6  is a graphic representation of the output voltage swing of the adaptive rail amplifier showing the positive and negative adaptive rail signals. 
         FIG. 7  is a graphic representation of the positive output signal swing of a 100 Hz sine wave showing the positive adaptive rail signal and positive pulsed rail control signal of  FIG. 3 ; 
         FIG. 8  is a simplified schematic diagram of a typical power amplifier block  14  of  FIG. 3 ; 
         FIG. 9  is a block diagram of a linear mode embodiment of the invention providing non-pulsed adaptive rails with reduced EMI emissions; 
         FIG. 10  is a schematic diagram of the positive charge pump circuit of  FIG. 9 ; 
         FIG. 11  is a schematic diagram of the negative charge pump circuit of  FIG. 9 ; 
         FIG. 12  is a graphic representation of the positive output signal swing of a 100 Hz sine wave showing the positive adaptive rail signal and positive adaptive rail control signal of  FIG. 9 ; 
         FIG. 13  is a block diagram of a single comparator charge pump tracking rail embodiment of the invention; 
         FIG. 14  is a block diagram of a switched dual comparator charge pump tracking rail embodiment of the invention; 
         FIG. 15  is a schematic diagram of the positive charge pump circuit of  FIG. 14 . 
         FIG. 16  is a schematic diagram of the negative charge pump circuit of  FIG. 14 ; 
         FIG. 17  is a schematic diagram of the positive adaptive rail circuit of  FIG. 14 ; 
         FIG. 18  is a schematic diagram of the negative adaptive rail circuit of  FIG. 14 ; and 
         FIG. 19  is a schematic diagram of an alternate or simplified embodiment of the adaptive rail power amplifier. 
     
    
    
     While the invention will be described in connection with preferred embodiments thereof, it will be understood that it is not intended to limit the invention to those embodiments or to the details of the construction or arrangement of parts illustrated in the accompanying drawings. 
     DETAILED DESCRIPTION 
     In the following description of the Figures, similar reference symbols designate corresponding structural parts or functional blocks. 
     The Prior Art Automotive Aftermarket Power Amplifier 
     Turning first to  FIG. 1 , a block diagram of the typical aftermarket automotive power amplifier is shown. The switch mode power supply  11  and the audio power amplifier  14  are contained in one chassis or unit. A 12 volt battery  10  provides the input power to the switch mode power supply  11  via a +12 volt and ground connection. The switch mode power supply  11  converts the 12 volt battery voltage to a bipolar output voltage referenced to ground, providing a positive output voltage PS and a negative output voltage NS, which supplies bipolar power to the amplifier  14 . The audio amplifier  14  receives an audio input signal and amplifies this audio input signal to drive a speaker  100 . 
     Looking at  FIG. 2 , a simplified schematic of the switch mode power supply  11  of  FIG. 1  is shown. The switch mode power supply  11  receives its input power from the 12 volt battery  10 , which is connected to the center tap of a transformer T 1 . A pulse width modulator controller  13  controls the switching of high power MOSFET switching transistors Q 1 , Q 2 , Q 3  and Q 4 . The switching topology is called a “push-pull” converter because the transformer T 1  has a “center-tapped” primary. The center tap is permanently connected to the 12 volt car battery, typically via an LC filter (not shown), to reduce switching spikes in the battery voltage. The ends of the primary side of the transformer T 1  are each connected to a different paralleled pair of MOSFETs that switch them to ground in each conduction cycle. MOSFETs Q 1  and Q 2  are parallel connected to one primary end and MOSFETs Q 3  and Q 4  are parallel connected to the other primary end so as to provide higher switching current than would be possible with a single MOSFET transistor. The secondary side of the switch mode power supply transformer T 1  has its center tap connected to ground and the two outputs of the transformer T 1  are connected to switching diodes in a diode bridge D 1 . The diodes of the bridge D 1  must be extremely fast diodes and are typically discrete diodes in a TO220 package with a 10 amp minimum rating and at a suitable voltage rating based on the output voltage of the transformer. The output of the diode bridge D 1  is filtered by capacitors C+ and C− at nodes PS and NS, respectively, to provide a positive and negative 30 volt high current output capable of providing the required voltage and current to power an audio amplifier. The typical switching frequency of this type of switch mode power supply is between 25 KHz and 100 KHz, which reduces the size of both the transformer T 1  and the output filter capacitors C+ and C. Due to the high switching frequency and the large current pulses generated when switching a high current square wave at high frequencies, this design suffers from high EMI/RFI emissions. When the audio power amplifier  14  is delivering large amounts of current to the speaker load, the RFI emissions can actually reduce FM radio reception and, in some cases, cause complete loss of certain AM radio band frequencies. As a result, the automotive OEMs have avoided using this type of switch mode power supply/power amplifier technology. 
     Pulsed Adaptive Rail Power Amplifier for Automotive Use 
     Turning now to  FIG. 3 , a block diagram of a pulsed adaptive rail power amplifier according to the invention is shown. The power amplifier  14  receives both positive and negative power from charge pump circuits  40  and  41 , respectively. The input of the power amplifier  14  is biased at VREF  28 , which is ½ the auto battery voltage. The battery voltage will be between 12 volts without the engine running and 14.4 volts with the alternator charging the battery. This means that VREF  28  will track at ½ the battery voltage, keeping the output voltage swing of the amplifier  14  centered between the positive battery voltage and is ground. A comparator  24  compares the output voltage swing of the power amplifier  14  and the positive adaptive rail voltage  21  at the output of the positive charge pump circuit  40 . A resistor  26  provides hysteresis for the comparator  24  which determines in part the switching frequency of the adaptive pulsed rail  21  and provides a window of switching between transitions. By increasing the hysteresis, a larger transition time between switching points occurs, which reduces the switching transitions of the positive cycling waveform. When the output voltage swing of the power amplifier  14  approaches the battery voltage, the output of the comparator  24  changes from a high output to low. Due to the positive feedback of the hysteresis resistor  26 , the output of the positive charge pump circuit  40  will increase only a small voltage based on the amount of hysteresis. This small voltage increase will cause the comparator  24  to change state again to high. This slight positive/negative cycle will continue as long as the output signal swing of the amplifier  14  requires more voltage than the input battery voltage at B 1 . This will provide a series of output pulses at the comparator  24  which change in pulse width and frequency based on the current demand on the output of the positive charge pump circuit  40 . As the amplifier  14  starts to swing negative with respect to the input battery voltage, the positive adaptive rail  21  will reach a quiescent point and the amplifier  14  will now draw its power from the battery voltage available at B 1 . The voltage difference between the output of the amplifier  14  and the adaptive rail voltage  21  is further defined by the rail offset voltage  27  applied to the negative input of the positive rail comparator  24 . This offset is adjusted to a level required to provide sufficient voltage for the output stage of the amplifier  14  to drive the load, as shown the speaker  100 , but also keeps the voltage across the output devices of the power amplifier  14  low so as to keep dissipated heat to a minimum. The output voltage of the positive adaptive rail  21  of the positive charge pump circuit  40  will track the output voltage swing of the amplifier  14  within the predefined offset voltage set by the rail offset  27  regardless of any drop in stored charge in the capacitors C 1 P and C 2 P of the positive charge pump circuit  40  (seen in  FIG. 4 ) due to the fact that the system operates within a feedback loop. The system compares the output signal of the power amplifier  14  plus the offset voltage  27  with the positive rail voltage  21  and switches the positive charge pump circuit  40  as required to keep the positive rail voltage  21  within the specified offset regardless of any loss in the stored charge of the capacitors C 1 P and C 2 P in the charge pump circuit  40  (see  FIG. 4 ). As mentioned above, when the output voltage swing of the amplifier  14  drops below the battery voltage by an amount equal to the offset voltage  27 , the positive rail voltage  21  returns back to the voltage of the battery. Thus, the positive charge pump circuit  40  allows an increase above the battery voltage up to two times the battery voltage as required to track the output voltage swing of the amplifier  14 . 
     When the output voltage swing of the amplifier  14  approaches ground, the output of the negative rail comparator  25  changes from high to low. Due to the positive feedback of its hysteresis resistor  29 , the output of the negative charge pump circuit  41  will decrease only a small voltage, based again on the amount of hysteresis. This small voltage decrease will cause the negative rail comparator  25  to change states again to high. This slight positive/negative cycle will continue, as long as the output signal swing of the power amplifier  14  requires more voltage than the ground side of the battery. This will provide a series of output pulses at the negative rail comparator  25  changing in pulse width and frequency based on the current demand on the output of the negative charge pump circuit  41 . As the power amplifier  14  starts to swing positive with respect to the battery ground, the negative adaptive rail  23  will reach a quiescent point and the power amplifier  14  will now draw its power from the battery ground. The voltage difference between the output of the power amplifier  14  and the negative adaptive rail  23  is also further defined by the rail offset  27  applied to the negative input of the negative rail comparator  25 . The system compares the output signal of the power amplifier  14  with the negative rail voltage  23  plus the offset voltage  27  and switches the negative charge pump circuit  41  as required to keep the negative rail voltage  23  within the specified offset regardless of any loss in the stored charge of the capacitors C 1 N and C 2 N in the negative charge pump circuit  41  (see  FIG. 5 ). 
     Both charge pump circuits  40  and  41  receive the battery voltage applied at B 1  and at G 1  the battery ground. The charge pump circuits  40  and  41  are capable of providing large amounts of output current at the positive adaptive rail  21  and the negative adaptive rail  23 . This allows the amplifier  14  to swing nearly three times the typical output voltage that would otherwise be available without the added voltage increase from charge pump circuits  40  and  41 . 
     The Charge Pump Circuits 
     Looking now at  FIG. 4 , a partial block, partial schematic diagram of the positive charge pump circuit  40  of  FIG. 3  is shown. A positive dead zone circuit  30 P receives the positive adaptive rail control signal  20  from its comparator  24 . The dead zone circuit  30 P provides an output signal with a minimum of 10 microseconds dead time between the negative going signal for the P-channel drive  31 P and the negative going signal for the N-channel drive  32 P. This avoids having the P-channel MOSFET  41 P and the N-channel MOSFET  42 P turned on at the same time, avoiding overheating of the switching devices or the device destruction which would occur if both MOSFETs  41 P and  42 P were on at the same time. The dead zone circuit  30 P also provides a 10 microseconds dead time when the positive adaptive rail control signal  20  goes positive. This will ensure that the N-channel power MOSFET transistor  42 P is completely off before switching on the P-channel power MOSFET transistor  41 P. When the positive charge pump circuit  40  is powered on, the N-channel MOSFET transistor  42 P will be turned on, connecting the negative plate of the first positive capacitor C 1 P to ground. This will charge the first positive rail capacitor C 1 P through its schottky diode DIP connected to the B 1  terminal, which is connected to the 12 volt battery or other power source. The first positive rail capacitor C 1 P is the flying capacitor of the positive charge pump circuit  40 , which transfers its charge through another schottky diode D 2 P into a second positive rail capacitor C 2 P, providing the output voltage  21  of the positive adaptive pulsed rail  21 . Both diodes D 1 P and D 2 P are schottky diodes which provide low forward voltage drop and fast switching response. The battery voltage at B 1  is also fed directly to the adaptive positive rail  21  through a third diode D 3 P. When the positive adaptive rail control signal  20  goes negative, the N-channel power MOSFET transistor  42 P switches off and the P-channel power MOSFET transistor  41 P switches on, connecting the negative plate of the flying capacitor C 1 P to the B 1  terminal, allowing forward conduction of the schottky diode D 2 P and transferring voltage from the flying capacitor OP to the positive pulsed rail  21 . The source connection of the P-channel power MOSFET transistor  41 P and the source connection of the N-channel power MOSFET transistor  42 P connect to the B 1  terminal and ground connection through ferrite beads L 1 P and L 2 P, respectively, which reduce the high speed switching transients reducing RFI emissions of the circuit. The second positive rail capacitor C 2 P is typically between 100 uF and 1000 uF for subwoofer applications which allows a fast release time for the positive adaptive rail  21 . When the positive adaptive rail control signal  20  goes high, the P-channel MOSFET transistor  41 P turns off and the N-channel MOSFET transistor  42 P turns on again, charging the positive rail flying capacitor C 1 P. 
       FIG. 5  is a partial block, partial schematic diagram of the negative charge pump circuit  41  of  FIG. 3 . Its dead zone circuit  30 N receives the negative adaptive rail control signal  22  from its comparator  25 . The negative dead zone circuit  30 N provides an output signal with a minimum of 10 microseconds dead time between the negative going signal for the P-channel drive  31 N and the negative going signal for the N-channel drive  32 N. This avoids having the P-channel MOSFET  41 N and the N-channel MOSFET  42 N turned on at the same time, avoiding overheating of the switching devices and the device destruction which would occur if both negative charge pump MOSFETs  41 N and  42 N were on at the same time. The negative dead zone circuit  30 N also provides a 10 microseconds dead time when the negative adaptive rail control signal  22  goes negative. This will ensure that the N-channel power MOSFET transistor  42 N is completely off before switching on the P-channel power MOSFET transistor  41 N. When the negative charge pump circuit  41  is powered on, the P-channel MOSFET transistor  41 N will be turned on, connecting the positive plate of its flying capacitor C 1 N to the terminal B 1 . This will cause the negative rail flying capacitor C 1 N to charge in cooperation with a first negative rail schottky diode D 1 N connected to the ground terminal G 1 , which is connected to the 12 volt battery ground or other power source. The flying capacitor C 1 N of the negative charge pump circuit  41  transfers its charge through another schottky diode D 2 N into a second capacitor C 2 N, providing the output voltage  23  of the negative adaptive pulsed rail  23 . Both diodes D 1 N and D 2 N are schottky diodes which provide low forward voltage drop and fast switching response. The battery ground is also fed directly to the adaptive negative rail  23  through a third diode D 3 N. When the positive adaptive rail control signal  22  goes positive, the P-channel power MOSFET transistor  41 N switches off and the N-channel power MOSFET transistor  42 N switches on, connecting the positive plate of the negative rail flying capacitor C 1 N to the ground terminal and allowing forward conduction of its schottky diode D 2 N, transferring voltage from the flying capacitor C 1 N to the negative rail  23 . The source connection of the P-channel power MOSFET transistor  41 N and the source connection of the N-channel power MOSFET transistor  42 P connect to the B 1  terminal and ground connection through ferrite beads L 1 P and L 2 P and L 1 N and L 2 N, respectively, which reduce the high speed switching transients and reduce RFI emissions of the circuit. The second negative rail capacitor C 2 N is typically between 100 uF and 1000 uF for subwoofer applications which allows a fast release time for the negative adaptive rail  23 . When the negative adaptive rail control signal  22  goes low, the P-channel MOSFET transistor  41 P turns off and the N-channel MOSFET transistor  42 N turns on, again charging the negative rail flying capacitor C 1 N. 
       FIG. 6  is a graph of the output voltage swing of the adaptive rail amplifier showing the power amplifier output signal and the positive and negative adaptive rail signals  21  and  23 . The amplifier output signal starts at VREF  28 , which is shown in this graph at  7  volts and is based on a battery supply voltage of 14 volts. This will allow the amplifier output signal swing to be centered between the 14 volt battery voltage and battery ground at 0 volts DC. When the output signal swing goes positive by more than battery voltage minus the preset rail offset voltage  27 , the positive supply rail increases with a series of positive going pulses, producing the positive adaptive rail  21 . The positive adaptive rail  21  will track the output signal offset positive by the rail offset  27 . When the output signal drops below the 14 volt battery voltage plus the pre-defined rail offset  27 , the positive rail returns back to the battery voltage at  14  volts. As the output signal swings negative and approaches the negative rail at 0 volts DC by more than the predefined rail offset  27 , the negative rail increases negative with a series of negative going pulses, producing the negative adaptive rail  21 . 
       FIG. 7  is a graph of the positive output signal swing of a 100 Hz sine wave showing the adaptive rail signal  21  and the positive pulsed rail control signal  20  of  FIG. 3 .  FIG. 7  shows in greater detail only the positive going portion of the signal shown in  FIG. 6  and further includes the positive pulsed rail control signal  20  in  FIG. 3 . As the output signal swings positive by an amount equal to the positive battery voltage  14  minus the predefined rail offset, the positive pulsed rail control starts to generate a series of pulses, which changes in pulse width depending on the amount of output current required to source the positive supply voltage to the power amplifier  14  in  FIG. 3 . The negative going pulses get wider as the output signal swings more positive and the demand for output current increases. When the signal starts to swing negative from the maximum peak swing, the output pulses change in both frequency and width until the output signal swings negative by more than the predefined rail offset  27  with respect to the 14 volt battery voltage. The positive pulsed rail control signal  20  stops changing and provides a positive signal allowing the flying capacitor C 1 P in the positive charge pump circuit  40  of  FIG. 4  to fully charge. The negative going signal will produce a similar but negative response from the negative adaptive rail control signal  22  and the negative charge pump circuit  41  shown in  FIG. 3 . 
     Typical Power Amplifier 
     Turning now to  FIG. 8 , a simplified schematic diagram of a typical power amplifier  14  of  FIG. 3  is shown. The power amplifier  14  includes a preamplifier/pre-drive circuit P 1  and a bipolar output stage including transistors Q 5  and Q 6 . The preamplifier P 1  receives a positive input signal via one resistor R 3  plus VREF  28  bias via another resistor R 4 . VREF  28  is also applied via a resistor R 1  to the negative input of the preamp/pre-drive circuit P 1 . The preamp circuit P 1  receives its power directly from the battery voltage applied at B 1  and battery ground. The pre-drive output signal drives the base of both transistors Q 5  and Q 6  which receive the positive adaptive rail voltage  21  and the negative adaptive rail voltage  23 , respectively. Amplifier negative feedback is provided via another resistor R 2 . 
     Many possible amplifier configurations will operate with the disclosed invention, including Class A/B, Class B, Class D and output stages utilizing bipolar or MOSFET transistors. As shown, the preamp/pre-drive circuit is connected directly to the battery voltage, but this stage of the power amplifier  14  can also be configured to connect to the adaptive rails  21  and  22 . If a single amplifier  14  is used, as in  FIG. 8 , the output speaker  100  will need to be decoupled from the amplifier by use of a large value capacitor (not shown) so as to eliminate the VREF bias from the output signal applied to the speaker  100 . The invention can also be implemented with dual amplifiers in bridge mode, effectively doubling the output voltage swing across the speaker load and eliminating the need for output decoupling capacitors. 
     While the above-disclosed embodiments of the invention provide excellent performance and improved manufacturing costs over the prior art, it may be desirable to eliminate all of the switching transients for use in certain automotive applications so as to eliminate virtually all of the RFI/EMI emissions of the amplification system. As can be seen with reference to  FIG. 3 , the system operates with a large gain in the feedback loop due to the comparators  24  and  25 . The invention can also be implemented with reduced loop gain, allowing more linear operation of the adaptive rails and eliminating the high frequency switching transients of the pulsed rails. The main tradeoff of this embodiment is that the MOSFETs that drive the positive adaptive rail and negative adaptive rail will see a slight increase in dissipated heat due to using them in a linear mode of operation. 
     Linear Mode Adaptive Rail Power Amplifier 
     In  FIG. 9 , a linear mode embodiment of the invention provides non-pulsed adaptive rails with reduced EMI emissions. The  FIG. 9  circuit is the same basic circuit as shown in  FIG. 3  with reduced feedback loop gain by changing the comparators  24  and  25  shown in  FIG. 3  to differential amplifiers  24  and  25 , as shown in  FIG. 9 . The power amplifier  14  receives both positive and negative power directly from the battery due to the schottky diodes D 3 P and D 3 N and also to the positive and negative charge pump circuits  40  and  41 , respectively. The amplifier input is biased at VREF  28 , which is ½ the auto battery voltage. As described above, the battery voltage will be between 12 volts without the engine running and 14.4 volts with the alternator charging the battery. This means that VREF  28  will track at ½ the battery voltage, keeping the output voltage swing of the amplifier  14  centered between the positive battery voltage and ground. A differential amplifier  24  provides an output signal which is the difference between the positive adaptive rail voltage  21  applied to the negative input via one resistor R 6  and the amplifier output signal applied to the positive input by another resistor R 7 . A third resistor R 5  provides negative feedback for the differential amplifier  24  and a fourth resistor R 8  provides positive rail offset  27 P applied to the positive input. The rail offset voltage  27 P applied to the fourth resistor R 8  is positive, typically 1.5 volts, and provides the proper offset required for the positive adaptive rail  21  to track the output signal. The gain of the differential amplifier  24  is selected to be less than one and typically on the order of 0.3 in order to avoid clipping the differential amplifier with a full scale output signal at the output of the amplifier  14 . If the differential amplifier  24  clips prior to detecting the maximum output signal swing, the positive adaptive rail control signal  20  will not provide proper rail tracking. The differential output signal  20  is fed to the input of the positive charge pump circuit  40 . The linear embodiment of the invention also operates in a feedback loop by taking the difference between the output amplifier&#39;s signal plus the offset voltage  27 P and the positive rail voltage  21  and controls the output voltage of the positive charge pump circuit  40  as required to keep the rail voltage  21  within the specified offset regardless of any loss in the stored charge of the capacitors C 1 P and C 2 P in the charge pump circuit  40  (See  FIG. 10 ). 
     Another differential amplifier  25  provides an output signal which is the difference between the negative adaptive rail  23  applied to the negative input via one resistor R 11  and the amplifier output signal applied to the positive input via another resistor R 10 . A third resistor R 12  provides negative feedback for the differential amplifier  25  and a fourth resistor R 9  provides negative rail offset  27 N applied to the positive input. The rail offset voltage  27 N applied to the fourth resistor R 9  is negative, typically −1.5 volts, and provides the proper offset required for the negative adaptive rail  23  to track the output signal. The gain of the differential amplifier  25  is selected to be less than one and typically on the order of 0.3 in order to avoid clipping the differential amplifier  25  with a full scale output signal at the output of the amplifier  14 . If the differential amplifier  25  clips prior to detecting the maximum output signal swing, the negative adaptive rail control signal  22  will not provide proper rail tracking. The differential output signal  22  is fed to the input of the negative charge pump circuit  41 . 
       FIG. 10  is a schematic diagram of the positive charge pump circuit  40  of  FIG. 9 . The positive charge pump circuit  40  receives the battery voltage at B 1  and ground and battery voltage applied to B 1  is fed to the positive adaptive rail  21  via a schottky diode D 3 P. The positive adaptive rail control  20  from the differential amplifier  24  of  FIG. 9  is applied to a resistor R 14  and to the positive input of a positive charge pump comparator U 1 . The positive charge pump comparator U 1  is connected to ½ battery voltage VREF and the output will be low when the output signal from the power amplifier  14  of  FIG. 9  is less than the battery voltage minus the rail offset  27 P applied to the differential amplifier  24 , also seen in  FIG. 9 . Continuing to look at  FIG. 10 , a first transistor Q 8  operates as a switch to provide gate drive for the N-channel power MOSFET transistor  42 P. The emitter of the transistor Q 8  is connected to the battery voltage B 1  and its collector is connected to ground through a resistor R 16 . The output of the positive charge pump comparator U 1  is connected via a 6 volt zener diode ZD 1  to the transistorbase drive resistor R 15 . When the positive charge pump comparator U 1  is low, the transistor Q 8  is turned on, providing gate drive and turning on the N-channel power MOSFET transistor  42 P, connecting the negative plate of the positive rail flying capacitor C 1 P to ground. A ferrite bead L 2 P is connected in series with the source connection of the N-channel MOSFET transistor  42 P in order to reduce the switching transient from the N-channel power MOSFET transistor  42 P. This allows the positive rail flying capacitor C 1 P to charge through the schottky diode D 1 P. When the positive adaptive rail control  20  goes positive by more than VREF, the positive charge pump comparator U 1  will switch high, turning the transistor Q 8  off and turning off the N-channel MOSFET  42 P. When the positive adaptive rail  20  goes high by more than VREF plus the VBE of the transistor Q 7 , the transistor Q 7  will start to conduct, providing gate drive to the P-channel power MOSFET transistor  41 P. If the positive rail flying capacitor C 1 P is fully charged, the P-channel power MOSFET transistor  41 P will operate in its linear region pulling the negative plate of its flying capacitor C 1 P positive. As the P-channel MOSFET transistor  41 P starts to conduct, the gate drive signal supplied from the output of the differential amplifier  24  in  FIG. 9  will be reduced and will thus provide a linear voltage increase that will track the output amplifier signal swing plus the rail offset voltage  27 P. As the signal swing drops below the battery voltage plus the rail offset voltage, the P-channel power MOSFET transistor  41 P will stop conducting. When the differential amplifier  24  output signal swings below VREF, the positive charge pump comparator U 1  will switch low, turning on the N-channel power MOSFET transistor  42 P, again providing charging for the positive rail flying capacitor C 1 P. The dead zone operation is provided due to the fact that the comparator U 1  switches at VREF and the transistor Q 7  starts to conduct only when the positive adaptive rail control signal  20  increases above VREF by one VBE. This ensures that both N-channel and P-channel power MOSFET transistors  41 P and  42 P cannot conduct at the same time. 
     In  FIG. 11 , the negative charge pump circuit  41  of  FIG. 9  receives battery voltage at B 1  and ground. Battery ground is fed to the negative adaptive rail  23  via the schottky diode D 3 N. The negative adaptive rail control  22  from the differential amplifier  25  of  FIG. 9  is applied to R 19  and to the positive input of a negative charge pump comparator U 2 . The negative charge pump comparator U 2  negative input is connected to ½ battery voltage VREF and the output will be high when the output signal from the amplifier  14  of  FIG. 9  is greater than the battery ground plus the rail offset  27 N applied to the differential amplifier  25  also of  FIG. 9 . The output of the negative charge pump comparator U 2  is connected to a 6 volt zener diode ZD 2  and to the base drive resistor R 17 . Transistor Q 9  operates as a switch and will switch off when the negative charge pump comparator U 2  is high. The emitter of the transistor Q 9  is connected to B 1  battery voltage and its collector is connected to ground through a resistor R 18 . When Q 9  is switched off, the P-Channel power MOSFET transistor  41 N is turned on due to a resistor R 18  pulling the P-Channel gate drive  31 N low. When P-Channel power MOSFET transistor  41 N is switched on, the positive plate of the negative rail flying capacitor C 1 N is tied to B 1  providing ch, transistor Q 9  is turned on eliminating gate drive  31 N and turning off the p-channel power MOSFET transistor  41 N. A ferrite bead L 1 N is connected in series with the source connection of the MOSFET transistor  41 N in order to reduce the switching transient from the power MOSFET transistor  41 N. When the negative adaptive rail  22  goes negative by more than VREF, the negative charge pump comparator U 2  will switch low, turning the transistor Q 9  on and turning off the P-channel MOSFET  41 N. When the negative adaptive rail control  22  is low by more than VREF plus the VBE of the other transistor Q 10 , the transistor Q 10  will start to conduct, providing gate drive to the N-channel power MOSFET transistor  42 N. If the negative rail flying capacitor C 1 N is fully charged, the N-channel power MOSFET transistor  42 N will operate in its linear region, pulling the positive plate of the negative rail flying capacitor C 1 N negative. As the MOSFET transistor  42 N starts to conduct, the gate drive signal supplied from the output of the differential amplifier  25  in  FIG. 9  will be reduced and will thus provide a linear negative voltage increase that will track the output amplifier signal swing plus the rail offset voltage  27 N from  FIG. 9 . As the signal swing goes above battery ground plus the rail offset voltage, the N-channel power MOSFET transistor  42 N will stop conducting. When the output signal of the differential amplifier  25  swings above VREF, the negative charge pump comparator U 2  will switch high, turning on the P-channel power MOSFET transistor  41 N again, providing charging for the negative rail flying capacitor C 1 N. Dead zone operation is provided due to the fact that the negative rail comparator U 2  switches at VREF and the transistor Q 9  starts to conduct only when negative adaptive rail signal  22  increases below VREF by one VBE. This ensures that both the N-channel power MOSFET and the P-channel power MOSFET transistors cannot conduct at the same time. Both positive and negative adaptive rails will track the amplifier output voltage swing plus the predefined positive and negative rail offset voltage up to the point that the signal swing in either direction exceeds the stored capacitor voltage available in the flying capacitors C 1 P and C 1 N in the positive and negative charge pump circuits  40  and  41 . The linear embodiment shown in  FIG. 9  provides an easy to manufacture design with reduced parts count and complexity. 
       FIG. 12  compares the positive output signal swing of a 100 Hz sine wave with the positive adaptive rail signal  21  and the positive adaptive rail control signal  20  of  FIG. 9 .  FIG. 12  shows only the positive going portion of a signal from  FIG. 9  and further includes the positive adaptive rail control  20  in  FIG. 9 . As the output signal swings positive by an amount equal to the positive battery voltage  14  minus the predefined rail offset, the positive adaptive rail control  20  starts to increase in voltage. The positive adaptive rail  21  linearly tracks the output signal plus a rail offset determined by the rail offset  27 P in  FIG. 9 . When the signal starts to swing negative from the maximum peak swing, the output voltage continues to track the output signal until the output signal drops below the battery voltage, shown as 14 volts plus the rail offset voltage. The output of the difference is amplifier  24  in  FIG. 9 , which produces positive adaptive rail control  20  shown in  FIG. 12 , will remain constant as the adaptive rail tracks the output signal so as to maintain a constant linear rail offset which tracks the output signal. The negative going signal will produce similar negative response from the negative adaptive rail control  22  and negative charge pump circuit  41  shown in  FIG. 9 . 
     Single Comparator Switched Charge Pump Tracking Rail 
     Referring now to  FIG. 13 , a single comparator switched charge pump tracking rail embodiment of the invention is shown.  FIG. 13  uses the same reference designations as in  FIG. 9  where similar circuit operation is disclosed. The power amplifier  14  output is connected to the positive input of the comparator  44  and the negative input of the comparator  44  is connected to VREF, which is ½ supply voltage. The comparator  44  provides a zero crossing detector and may include positive hysteresis so as to avoid comparator switching due to noise or very low level signals. When the output signal from the power amplifier  14  swings positive above VREF, the output  45  of the comparator  44  switches high and when the output signal of the power amplifier  14  swings negative below VREF, the output  45  of the comparator  44  switches low. The comparator  44  generates a control signal  45  to control the positive and negative charge pump circuits  40  and  41 , which generates a positive 2× voltage at the output  42  of the positive charge pump circuit  40  and a negative 2× output voltage at the output  43  of the negative charge pump circuit  41 . The charge pump circuits  40  and  41  are clocked by the audio signal even when the audio signal is low and does not require any additional output swing beyond the limits of the supplied battery voltage. Upon power up, the flying capacitors C 1 P and C 1 N in both charge pump circuits  40  and  41  are charged up to the battery voltage. With each positive swing of the output signal of the power amplifier  14 , the internal flying capacitor C 1 P of the positive charge pump circuit  40  will transfer a charge to an internal output storage capacitor C 1 P within the positive charge pump circuit  40 , thereby increasing the positive charge pump output voltage  42 . With each negative swing of the output signal of the power amplifier  14 , the internal flying capacitor C 1 N of the negative charge pump circuit  41  will transfer a charge to an internal output storage capacitor C 2 N within the negative charge pump circuit  41 , thereby increasing the negative charge pump output voltage  43 . The positive 2× voltage  42  is fed to the input of the positive adaptive rail circuit  50 . The positive adaptive rail circuit  50  is controlled via the control signal  20  from the differential amplifier  24  and provides a positive adaptive rail output  21  which increases the positive power supply voltage above the battery voltage B 1  which is connected to a schottky diode D 3 P. The nominal rail voltage  21  will be the same as the battery voltage B 1  less the forward drop of the schottky diode D 3 P and will increase up to two times the battery voltage less any capacitor voltage sag that may occur in the positive charge pump circuit  40 . The negative 2× voltage  43  is fed to the input of the negative adaptive rail circuit  51 . The negative adaptive rail circuit  51  is controlled via the control signal  22  from the differential amplifier  25  and provides a negative adaptive rail output  23  which increases the negative power supply voltage below the battery ground connected to a schottky diode D 3 N. The nominal rail voltage  23  will be the same as battery ground less the forward drop of the schottky diode D 3 N and will increase negative up to two times the battery voltage less any capacitor voltage sag that may occur in the negative charge pump circuit  41 . When the output voltage swing of the power amplifier  14  approaches the positive rail voltage  21  minus the rail offset  27 P, the positive adaptive rail circuit  50  will become active and increase the positive rail  21  so as to keep the positive rail above the output voltage swing by an amount equal to the rail offset  27 P. When the output voltage swing of the power amplifier  14  approaches the negative rail voltage  23  minus the rail offset  27 N, the negative adaptive rail circuit  51  will become active and increase the negative rail voltage  23  so as to keep the negative rail below the amplifier  14  output voltage swing by an amount equal to the rail offset  27 N. 
     Looking at  FIG. 14 , a modification to the embodiment of  FIG. 13  can be made using two separate comparators  44 P and  44 N in place of the single comparator  44 . That would allow the flying capacitors C 1 P and C 1 N of both the positive and the negative charge pump circuits  40  and  41  to charge on power up or idle (no audio signal present). One comparator  44 P would compare the amplifier output signal swing with a positive reference voltage  28 P to control the positive charge pump circuit  40  and the second comparator  44 N would compare the amplifier output signal swing with a negative reference voltage  28 N to control the negative charge pump circuit  41 . This modification would allow the flying capacitors C 1 P and C 1 N in the positive and negative charge pump circuits  40  and  41  to both fully charge when the system is switched on or when no audio is present. Other modifications will become apparent to the skilled artisan. 
     Referring now to  FIG. 15 , the positive charge pump circuit  40  of the embodiment of  FIG. 14  is the same circuit as shown in  FIG. 10  without the shottky diode D 3 P and with changes in value for the capacitors C 1 P and C 2 P to optimize performance for this embodiment. As a result, the detailed description of  FIG. 15  is the same as the description given for  FIG. 10 . After a few switching cycles of the comparator  44 P of  FIG. 14 , the second positive charge pump capacitor C 2 P will charge up to a voltage equal to two times the battery voltage minus the forward diode drop of the schottky diodes D 1 P and D 2 P. There is no discharge path for the stored charge in the capacitor C 2 P until the positive adaptive rail circuit  50  of  FIG. 14  becomes active, so the stored charge will remain at this peak until the positive adaptive rail circuit  50  becomes active. When the positive adaptive rail circuit  50  of  FIG. 14  becomes active, the positive charge pump comparator  44 P will be switched positive, turning on the P-channel MOSFET  41 P of  FIG. 15 . This allows the stored charge in both positive charge pump capacitors C 1 P and C 2 P to provide current to increase the positive adaptive rail  21  of  FIG. 14 . This provides additional benefits by increasing the maximum stored charge available to the positive adaptive rail circuit  50  and decreasing the inrush current required to charge the single positive rail flying capacitor C 1 P on startup. The capacitor C 2 P and the diode D 2 P could be omitted, reducing the maximum available stored charge and providing a switched output voltage at  42  when the positive charge pump comparator  44 P of  FIG. 14  switches positive. The output signal  42  would connect to the cathode side of the schottky diode D 1 P in this configuration. The main advantage of this modification is reduced circuit complexity. 
     Referring now to  FIG. 16 , the negative charge pump circuit  41  of the embodiment of  FIG. 14  is shown.  FIG. 16  is the same circuit as shown in  FIG. 11  without the schottky diode D 3 N and with changes in value for the capacitors C 1 N and C 2 N to optimize performance for this embodiment. As a result, the detailed description of  FIG. 16  is the same as the description given for  FIG. 11 . After a few switching cycles of the comparator  44 N of  FIG. 14 , the second negative charge pump capacitor C 2 N will charge up to a voltage equal to minus two times the battery voltage minus the forward diode drop of the schottky diodes  131 N and D 2 N. There is no discharge path for the stored charge in the capacitor C 2 N until the negative adaptive rail circuit  51  of  FIG. 14  becomes active, so the stored charge will remain at this negative peak until the negative adaptive rail circuit  51  becomes active. When the negative adaptive rail circuit  51  of  FIG. 14  becomes active, the negative charge pump comparator  44 N will be switched negative. Looking again at  FIG. 16 , this turns on the N-channel MOSFET  42 N and allows the stored charge in both negative charge pump capacitors C 1 N and C 2 N to provide current to increase the negative adaptive rail  23  of  FIG. 14 . This provides additional benefits by increasing the maximum stored charge available to the negative adaptive rail circuit  51  and decreasing the inrush current required to charge the single capacitor C 1 N on startup. The capacitor C 2 N and diode D 2 N could be omitted, reducing the maximum available stored charge and providing a switched output voltage at  43  when the negative charge pump comparator  44 N of  FIG. 14  switches negative. The output signal  43  would connect to the anode side of the schottky diode DIN in this configuration. As mentioned above, the main advantage of this modification is reduced circuit complexity. 
       FIGS. 17 and 18  show the positive and negative adaptive rail circuits  50  and  51  of  FIG. 14 , respectively. As seen in  FIG. 17 , the positive adaptive rail circuit  50  includes the P-channel power MOSFET transistor  52  which receives the input signal  42  and provides a variable output voltage  21 . The P-channel power MOSFET  52  is controlled by a small signal transistor Q 7 . The transistor Q 7  receives the control signal  20  through a resistor R 14  to the base connection of the transistor Q 7 . The emitter of the transistor Q 7  is tied to VREF ½ supply and the collector of the transistor Q 7  is connected to the gate of the P-channel MOSFET  52  through a gate resistor and also to a resistor R 13  which is tied to input signal  42 . In operation, when a positive going control signal  20  appears, the transistor Q 7  becomes conductive, providing gate drive to the P-channel MOSFET  52  which in turn increases the output voltage  21 , providing an increase above the battery voltage seen at the power amplifier  14  of  FIG. 14 . By connecting the resistor R 13  to the input signal  42 , the power P-channel MOSFET  52  does not become active until the transistor Q 7  starts to conduct. The differential amplifier  24  of  FIG. 14  provides the control signal to the base of the transistor Q 7 . The output of the differential amplifier  24  is a voltage equal to the difference between the power amplifier  14  output voltage swing and the positive adaptive rail  21  plus the rail offset voltage  27 P. This will keep the positive adaptive rail voltage  21  above the output signal swing by an amount equal to the rail offset  27 P at all times up to the point where the output signal swings beyond the available voltage at  42  and the output clips of the power amplifier  14 . 
     Turning to  FIG. 18 , the negative adaptive rail circuit  51  includes the N-channel power MOSFET transistor  53  which receives the input signal  43  and provides a variable output voltage  23 . The N-channel power MOSFET  53  is controlled by a small signal transistor Q 10 . The transistor Q 10  receives the control signal  22  through a resistor R 19  to the base connection of the transistor Q 10 . The emitter of the transistor Q 10  is tied to VREF ½ supply and the collector of the transistor Q 10  is connected to the gate of the N-channel MOSFET  53  through a gate resistor and also to a resistor R 20  which is tied to the input signal  43 . In operation, when a negative going control signal  22  appears, the transistor Q 10  becomes conductive, providing gate drive to the N-channel MOSFET  53  which in turn increases the negative output voltage  23 , providing an increase below the battery ground G 1  seen at the power amplifier  14  of  FIG. 14 . By connecting the resistor R 20  to the input signal  43 , the power N-channel MOSFET  53  does not become active until the transistor Q 10  starts to conduct. The differential amplifier  25  of  FIG. 14  provides the control signal to the base of the transistor Q 10 . The output of the differential amplifier  25  is a voltage equal to the difference between the output voltage swing of the power amplifier  14  and the negative adaptive rail  23  plus the rail offset voltage  27 N. This will keep the negative adaptive rail voltage  23  below the output signal swing by an amount equal to the rail offset  27 N at all times up to the point where the output signal swings negative beyond the available voltage at  43  and the output clips of the power amplifier  14 . 
     In operation, all embodiments of the invention will produce a signal swing of nearly 40 volts peak into an 8 ohm load which will deliver 100 watts and nearly 180 watts into a 4 ohm load. As previously noted, higher impedance 8 ohm and 4 ohm speakers have higher sensitivity ratings and will therefore provide higher output sound pressure levels than the lower impedance speakers of 2 or 1 ohms. The invention, therefore, also allows higher sound pressure levels with reduced current consumption. 
     Simplified Embodiment of the Adaptive Rail Power Amplifier 
     Turning now to  FIG. 19 , typical Battery voltage of 12 volts (14.4 volts with the alternator running) is applied at B 1  and Battery Ground is applied at G 1 . Power amplifier  14  receives a positive power supply voltage from Positive Adaptive Rail  21  and receives negative power supply voltage from Negative Adaptive Rail  23 . With no input signal, or a low level input signal, the voltage on Positive Adaptive Rail  21  will equal the Battery voltage less one diode drop from schottky diode D 1 P and the Negative Adaptive Rail  23  will equal the Battery ground less one diode drop from schottky diode D 1 N. A reference voltage of ½ B 1  (battery voltage) is applied as signal VREF as a bias reference at the input and also at nodes indicated as VREF. The VREF signal will track the battery and provide ½ battery reference voltage as the battery voltage fluctuates. The output of amplifier  14  will be centered at VREF half way between B 1 , the positive battery voltage, and G 1 , the ground side of the battery. The emitter of Q 1  is connected to node  21 , which is equal to B 1  minus the forward diode drop of schottky diode D 1 P. The base of Q 1  is connected to 5.1K resistor R 1  and to the cathode side of zener diode D 2 . The anode side of D 2  is connected to the output of amplifier  14 . When the output swing of amplifier  14  is low, base current will flow in Q 1 , turning Q 1  on, which will in turn pull gate resistor R 2  positive ensuring that P-CHANNEL power MOSFET transistor will be turned off. This keeps P-CHANNEL power MOSFET transistor  41 P turned off until the output voltage swing of amplifier  14  exceeds the point where Q 2  switches off plus a predefined dead zone voltage. Q 2  emitter is tied to VREF and base resistor R 5  is tied to ground which will turn on Q 2  with no input signal thereby turning on N-CHANNEL power MOSFET transistor  42 P by pulling gate resistor R 7  positive causing storage capacitor C 1 P to charge through schottky diode D 1 P. When the output swing of amplifier  14  increases positive above VREF by more than one volt Q 2  will switch off pulling the gate of N-CHANNEL power MOSFET transistor  42 P low due to collector connected pull down resistor R 6 . Q 2 &#39;s base drive is a result of the divider voltage of R 5 /R 4 +R 5  times VOUT of power amplifier  14 . The VBE of Q 2  is effectively cancelled out by the forward drop of diode D 3  making the switch point of Q 2  as described above. With the values shown in  FIG. 1  and a battery voltage of 12 volts Q 2  will switch off when the output signal swing exceeds approximately 7 volts positive, one volt above VREF. As the output signal of amplifier  14  continues to swing positive approaching B 1 , Q 1  will stop conducting. When power amplifier  14  output signal approaches B 1 , minus the diode drop of schottky diode D 1 P, minus the VBE of transistor Q 1  and the zener voltage of zener diode D 2  transistor Q 1  will stop conducting. Zener diode D 2  will typically be a 1N5221 which has a 2.4 volt zener voltage. This will provide a dead zone of approximately 1.6 volts between the point where transistor Q 2  switches off and the point where transistor Q 1  will start to turn off allowing P-CHANNEL power MOSFET transistor  41 P to become active. The Source connection of P-CHANNEL power MOSFET transistor  41 P is connected directly to the B 1  positive battery connection. As power MOSFET transistor  41 P becomes active it will pull the negative plate of storage capacitor C 1 P positive towards the B 1  battery potential thereby increasing the Positive Adaptive Rail  21  supply voltage. As the Positive Adaptive Rail  21  increases, Q 1 &#39;s emitter voltage increases positive, which keeps Q 1  in a linear or non-saturated condition providing a positive adaptive power supply rail that will track the output signal by 3.4 volts positive above the output signal swing. If the output voltage swing exceeds the battery voltage plus the available stored charge in capacitor C 1 P, the positive output swing will start to clip. The Positive Adaptive Rail voltage will start to increase above the battery voltage when the output voltage swing increases positive by more than B 1  minus the forward drop of schottky diode D 1 P, minus the VBE of Q 1  minus the zener voltage of zener diode D 2 . With a 2.4 volt zener diode this offset will be approximately 3.4 volts. The offset voltage can be increased by changing zener diode D 2 , thus it will be apparent to the skilled artisan that increasing the zener voltage will increase the offset voltage by an equal amount. 
     The operation of the Negative Adaptive Rail will now be described. The emitter of Q 4  is connected to node  23 , which is equal to G 1  minus the forward diode drop of schottky diode D 1 N. The base of Q 4  is connected to 5.1K resistor R 14  and to the anode side of zener diode D 5 . The cathode side of D 5  is connected to the output of amplifier  14 . When the output voltage swing of amplifier  14  is low, base current will flow in Q 4 , turning Q 4  on, which will in turn pull gate resistor R 9  negative ensuring that P-CHANNEL power MOSFET transistor  42 N will be turned off. This keeps P-CHANNEL power MOSFET transistor  42 N turned off until the output voltage swing of amplifier  14  swings negative and exceeds the point where Q 3  switches off plus a predefined dead zone voltage. Q 3  emitter is tied to VREF and base resistor R 9  is tied to B 1  which will turn on Q 3  with no input signal thereby turning on N-CHANNEL power MOSFET transistor  41 N by pulling gate resistor R 10  positive causing storage capacitor C 1 N to charge through schottky diode D 1 N. When the output swing of amplifier  14  increases negative below VREF by more than one volt Q 3  will switch off pulling the gate of N-CHANNEL power MOSFET transistor  41 N high due to collector connected pull up resistor R 8 . Q 3 &#39;s base drive is a result of the divider voltage of R 9 /R 11 +R 9  times VOUT of power amplifier  14 . The VBE of Q 3  is effectively cancelled out by the forward drop of diode D 4  making the switch point of Q 3  as described above. With the values shown in  FIG. 1  and a battery voltage of 12 volts Q 3  will switch off when the output signal swing exceeds approximately 7 volts negative, one volt below VREF. As the output signal of amplifier  14  continues to swing negative approaching G 1 , Q 4  will stop conducting. When power amplifier  14  output signal approaches G 1 , minus the diode drop of schottky diode D 1 N, plus the VBE of transistor Q 4  and the zener voltage of zener diode D 5  transistor Q 3  will stop conducting. Zener diode D 5  will typically be a 1N5221 which has a 2.4 volt zener voltage. This will provide a dead zone of approximately 1.6 volts between the point where transistor Q 3  switches off and the point where transistor Q 4  will starts to turn off allowing N-CHANNEL power MOSFET transistor  42 N to become active. The Source connection of N-CHANNEL power MOSFET transistor  42 N is connected directly to the G 1  negative battery connection. As power MOSFET transistor  42 N becomes active it will pull the positive plate of storage capacitor C 1 N negative towards the G 1  battery ground potential thereby increasing the Negative Adaptive Rail  23  supply voltage. As the Positive Adaptive Rail  23  increases negative Q 4 &#39;s emitter voltage increases negative, which keeps Q 4  in a linear or non-saturated condition providing a negative adaptive power supply rail that will track the output signal by 3.4 volts negative below the output signal swing. If the output voltage swing exceeds the battery ground potential plus the available stored charge in capacitor C 1 N, the negative output swing will start to clip. The Negative Adaptive Rail voltage will start to increase below the battery ground potential when the output voltage swing increases negative by more than G 1  minus the forward drop of schottky diode DIN, minus the VBE of Q 4  and the zener voltage of zener diode D 5 . With a 2.4 volt zener diode this offset will be approximately 3.4 volts. The offset voltage can be increased by changing zener diode D 5 , thus it will be apparent to the skilled artisan that increasing the zener voltage will increase the negative offset voltage by an equal amount. 
     The circuit of  FIG. 19  will provide a positive and negative adaptive rail that will track the output signal by a predefined offset voltage and will charge the storage capacitors C 1 P and C 1 N when the power amplifier  14  output signal is low in amplitude or on power up when the amplifier output is zero. The circuit of  FIG. 19  will also provide accurate positive and negative adaptive tracking rails regardless of the amount of drop in voltage in the storage capacitors C 1 P and C 1 N. In operation the output signal swing of amplifier  14  and positive adaptive rail signal  21  will be virtually identical to that shown graphically in  FIG. 12  of the co-pending application. 
     It will be apparent to the skilled artisan that the circuit of  FIG. 19  can be implemented as a bridge mode design with one amplifier driving one side of the speaker in phase and a second channel of amplification driving the other side of the speaker out of phase, thereby providing twice the voltage swing and four times the output power. It will also be apparent to the skilled artisan that the circuit shown in  FIG. 19  will have advantages and can be implemented in a bipolar power supply design with a positive voltage, a ground reference and a negative voltage eliminating the need for a VREF reference bias voltage.