Abstract:
A method and apparatus receiving wireless signals and generating an output signal and diagnostic information therefrom is disclosed. Each of the plurality of wireless signals comprises video information and is characterized by a wireless frequency band, polarization, and apparent spatial direction, and the output signal comprises the video information of any the plurality of wireless signals, frequency translated to one of a plurality of stacked output signal frequency bands.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is related to U.S. application Ser. No. 12/195,256, filed on Aug. 20, 2008, by Robert F. Popoli, entitled “COMPUTATIONALLY EFFICIENT DESIGN FOR BROADCAST SATELLITE SINGLE WIRE AND/OR DIRECT DEMOD INTERFACE,” which application is incorporated by reference herein. 
     This application is a continuation of U.S. application Ser. No. 13/554,022, filed on Jul. 20, 2012, which is a continuation of U.S. application Ser. No. 12/683,270, filed on Jan. 6, 2010, now U.S. Pat. No. 8,229,383, which claims the benefit under 35 U.S.C. Section 119(e) of U.S. Provisional Application Ser. No. 61/142,865, filed on Jan. 6, 2009, by Robert F. Popoli, entitled “FREQUENCY DRIFT ESTIMATION FOR LOW COST OUTDOOR UNIT FREQUENCY CONVERSIONS AND SYSTEM DIAGNOSTICS,” which applications are incorporated by reference herein. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates generally to satellite video systems, and in particular, to a method, apparatus, and article of manufacture for frequency drift estimation in satellite outdoor unit systems. 
     2. Description of the Related Art 
     Satellite broadcasting of communications signals has become commonplace. Satellite distribution of commercial signals for use in television programming currently utilizes multiple feedhorns on a single Outdoor Unit (ODU) which supply signals to up to eight Integrated Receiver/Decoders (IRDs), e.g., set top boxes, on separate cables from a multiswitch. 
       FIG. 1  illustrates a typical satellite-based broadcast system of the related art. 
     System  100  uses signals sent from Satellite A (SatA)  102 , Satellite B (SatB)  104 , and Satellite C (SatC)  106  that are directly broadcast to an Outdoor Unit (ODU)  108  that is typically attached to the outside of a house  110 . ODU  108  receives these signals and sends the received signals to IRD  112 , which decodes the signals and separates the signals into viewer channels, which are then passed to monitor  114  for viewing by a user. There can be more than one satellite transmitting from each orbital location (slot). The orbital slots are typically designated by their longitude, so, for example, a satellite  102  located in the orbital slot at 101 degrees West Longitude (WL) is usually referred to as transmitting from “101.” 
     Satellite uplink signals  116  are transmitted by one or more uplink facilities  118  to the satellites  102 - 106  that are typically in geosynchronous orbit. Satellites  102 - 106  amplify and rebroadcast the uplink signals  116 , through transponders located on the satellite, as downlink signals  120 . Depending on the satellite  102 - 106  antenna pattern, the downlink signals  120  are directed towards geographic areas for reception by the ODU  108 . 
     Each satellite  102 - 106  broadcasts downlink signals  120  in typically thirty-two (32) different frequencies, which are licensed to various users for broadcasting of programming, which can be audio, video, or data signals, or any combination. These signals are typically located in the Ku-band of frequencies, i.e., 11-18 GHz, but can also be broadcast in the Ka-band of frequencies, i.e., 18-40 GHz, but typically 20-30 GHz. 
     Within ODU  108 , the downlink signals  120  are downconverted to lower frequencies using an oscillator and a mixer. Typically, the oscillator is a Dielectric Resonant Oscillator (DRO). If the DRO frequency drifts, the downconversion of the downlink signals  120  drifts as well, which makes processing of such downconverted signals more difficult. 
     Further, as satellites  102 - 106  broadcast additional services and additional channels to viewers, as well as additional satellite signals present in such bandwidths, it will be more efficient to monitor and, if necessary, correct drifts in the DRO frequency. 
     SUMMARY OF THE INVENTION 
     To minimize the limitations in the prior art, and to minimize other limitations that will become apparent upon reading and understanding the present specification, systems and devices for controlling frequency drift in satellite broadcast systems are presented herein. 
     A receiver antenna system for a direct broadcast satellite signal communications system in accordance with one or more embodiments of the present invention comprises an oscillator, a mixer, coupled to the oscillator, for converting satellite signals at a first frequency to signals at an intermediate frequency, an analog-to-digital (A/D) converter, coupled to the mixer, for receiving the signals at the intermediate frequency and for converting the signals at the intermediate frequency at near-real-time to a digital data stream, a Digital Signal Processor (DSP), coupled to the A/D converter, for processing the digital data stream, and a drift estimator, coupled to the DSP, the drift estimator determining a frequency drift of the oscillator, wherein the receiver antenna system corrects the frequency drift of the oscillator using the determined frequency drift. 
     Such a system further optionally comprises the drift estimator driving a digital mixer within the DSP to compensate for the determined frequency drift, an output of the drift estimator being fed back to the oscillator to control the frequency drift of the oscillator, an automatic gain control coupled between the mixer and the A/D converter, the A/D converter sampling the signals at the intermediate frequency at a speed greater than 1 gigasample per second, the satellite signals being transmitted in at least the Ku-band of frequencies, the satellite signals being further transmitted in at least the Ka-band of frequencies, a Digital-to Analog Converter (DAC), coupled to the DSP, and an output of the DAC being mixed with a second oscillator, such that an output of the receiver antenna system is set to a desired band on a single wire interface. 
     A system for distributing a plurality of satellite signals on a single interface in accordance with one or more embodiments of the present invention comprises an oscillator for down-converting the plurality of satellite signals to signals at an intermediate frequency, an Automatic Gain Controller (AGC) for gain controlling the signals at the intermediate frequency, an analog-to-digital (A/D) converter, coupled to the AGC, for receiving the signals at the intermediate frequency, wherein the A/D converter directly samples the signals at the intermediate frequency, and a Digital Signal Processor (DSP), coupled to the A/D converter, wherein a first output of the DSP is used to determine the intermediate frequency and a second output of the DSP is an input to the single interface. 
     Such a system further optionally comprises a Digital-to-Analog Converter (DAC), coupled to the second output of the DSP, the first output of the DSP determining the intermediate frequency by controlling a frequency of the oscillator, the first output of the DSP driving a compensatory frequency shift by controlling a digital mixer internal to the DSP, the oscillator being a Dielectric Resonance Oscillator (DRO), the plurality of satellite signals being transmitted in a plurality of frequency bands, the plurality of frequency bands comprising a Ka-band and a Ku-band, the DRO down-converting the Ka-band to at least a first intermediate frequency band and the DRO down-converting the Ku-band to a second intermediate frequency band as convenient for A/D conversion, the A/D converter sampling the signals at the intermediate frequency at a rate greater than the Nyquist rate for the signals at the intermediate frequency, state information from at least one of the AGC, the ADC, the DSP, and the DAC providing a diagnostic output for the system, and the diagnostic output comprising at least one of fault recognition, fault reporting, performance monitoring, installation aiding, and installation verification for the system. 
     Other features and advantages are inherent in the system disclosed or will become apparent to those skilled in the art from the following detailed description and its accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Referring now to the drawings in which like reference numbers represent corresponding parts throughout: 
         FIG. 1  illustrates a typical satellite-based broadcast system of the related art; 
         FIG. 2  illustrates a typical Single Wire Multiswitch (SWM) of the related art; 
         FIGS. 3A and 3B  illustrate related LNB modules and an LNB module in accordance with one or more embodiments of the present invention, respectively; 
         FIG. 4  illustrates an Analog-to-Digital subsystem functionality in accordance with one or more embodiments of the present invention; 
         FIG. 5  illustrates a block diagram of the Digital Single Wire Multiswitch (DSWM) Channelizer in accordance with one or more embodiments of the present invention; 
         FIG. 6  illustrates a coarse granularity channelizer in accordance with one or more embodiments of the present invention; 
         FIG. 7  illustrates an exemplary fine granularity channelizer embodiment in accordance with one or more embodiments of the present invention; and 
         FIG. 8  illustrates a block diagram of a frequency drift control system in accordance with one or more embodiments of the present invention. 
     
    
    
     DETAILED DESCRIPTION 
     In the following description, reference is made to the accompanying drawings which form a part hereof, and in which are shown, by way of illustration, several embodiments of the present invention. It is understood that other embodiments may be utilized and structural changes may be made without departing from the scope of the present invention. 
     Overview 
     Recent advances in high speed Analog to Digital (A/D) converters open up the possibility of direct A/D conversion of baseband or near baseband 500 MHz wide Satellite downlink signals. This allows for all-digital demultiplexing (demuxing) of one or more 500 MHz complexes or transponders and subsequent all-digital multiplexing (muxing) of a selected subset of these transponder channels onto a single wire interface for home distribution. With such a digital implementation, selected transponder baseband I/Q signals may be selected for direct demodulation, for subsequent home or multi-dwelling distribution over any suitable physical/network layer protocols. 
     By integrating the high-speed A/D hardware with computationally-efficient Digital Signal Processing (DSP) techniques, and adapting them specifically for the purposes of demuxing, muxing, and baseband I/Q conversion, one or more embodiments of the present invention provide significant commercial advantages over current analog designs. Further, simple brute force digital mimicry of known analog building blocks for demuxing/muxing/conversion processes are expected to lead to less commercially-viable designs. 
     No known previously proposed satellite home/MDU distribution design exists to create a practical all digital implementation of the demuxing, muxing, and direct digital demodulation (demod) interface. This specification proposes a number of embodiments of such designs which utilize computationally efficient techniques which allow for commercially viable all digital implementation of these functions. 
     The present invention allows for better overall performance of an embodiment of a Single Wire Multiswitch (SWM) architecture because the present invention allows for the distribution of more channels within the same bandwidth (i.e., single wire bandwidth) through tighter packing of channels. Distribution of additional channels allows for the support of additional IRDs, or the same number of IRDs with less wired bandwidth. Further, embodiments of the present invention allow for inexpensive provision of baseband I/Q signals for reduced cost integration of a significant portion of the current IRD. Further still, embodiments of the present invention allow for the ability to provide greater flexibility for future system designs, and the potential of a smaller footprint for the SWM into the current system. Embodiments of the present invention also allow for simpler integration and interfaces between the parts of the present system, by simplifying the interface of shared demodulation resources without expending any bandwidth. This simplification allows for future products, such as Home Gateway and Multi-Dwelling Unit (MDU) architectures, to become possible. 
     Digital SWM (DSWM) 
       FIG. 2  illustrates a typical Analog Single Wire Multiswitch. 
     Hardware advances have increased A/D sampling rates in excess of 1 Gigasample/second (1 GSPS) with a good Effective Number Of Bits (ENOB) and an adequate linearity performance figure. Advances in multi-rate Digital Signal Processing techniques coupled with nanometer Application-Specific Integrated Circuit (ASIC) processes allow applications with significant signal processing capabilities. These factors make it possible to make an all-digital replacement for the demux and mux functions of the Single Wire Multiswitch (SWM). Further, these technological advances allow for a baseband and/or near baseband digital I/Q interface for cost-effective integration of a significant portion of the IRD front end functionality. 
     One or more embodiments of the present invention provide a digital replacement for the hardware shown in  FIG. 2 .  FIG. 2  illustrates SWM  200 , where SWM  200  is fed by four composite signals  202 - 208 , specifically LNB 1   202 , LNB 2   204 , LNB 3   206 , and LNB 4   208 , which are produced by the LNB module  300  hardware shown in  FIG. 3A . 
     Each LNB signal  202 - 208 , as shown in  FIG. 3A , comprises multiple (e.g., three) stacked 500 MHz bandwidth signals, generated from downlink signals  120  and received at various orbital slots. Ku-band signals  302  are from satellites at the 119 WL slot, Ku-band signals  304  are from satellites at the 110 WL slot, Ka-band signals  306  are from satellites at the 102.8 (also referred to as 103) WL slot, Ku-band signals  308  are from satellites at the 101 WL slot, and Ka-band signals  302  are from satellites at the 99.2 (also referred to as 99) WL slot. Combinations of these signals, based on their polarization and transmission frequencies, are used to generate LNB signals  202 - 208 . 
     After down-conversion within module  300 , the Ku-band signals  302 ,  304 , and  308  are down-converted to an Intermediate Frequency (IF) band, 500 MHz wide, in the frequency range of 950-1450 MHz. The Ka-band signals  306  and  310  are down-converted into two different 500 MHz wide bandwidths, namely the 250-750 MHz bandwidth (known as Ka-LO IF band or Ka-B IF band) and the 1650-2150 MHz bandwidth (known as Ka-HI IF band or Ka-A IF band), and are combined in various combinations to form signals  202 - 208   
     The SWM  200  shown in  FIG. 2  selects any nine 40 MHz pieces of spectrum from LNB 1   202  through LNB 4   208  and stacks them to form a single composite signal called the channel stacked output  210 . The nine 40 MHz channels are typically located on 102 MHz centers and range from 974 MHz to 1790 MHz (i.e., the channels are at 974, 1076, 1178, 1280, 1382, 1484, 1586, 1688, and 1790 MHz, respectively). 
     Coarse Granularity Design 
       FIG. 3B  illustrates an embodiment an LNB structure corresponding to a Digital SWM Design in accordance with the present invention. 
     The SWM module  300  is modified into module  312 , where signals  302 - 310  are combined into different signals that are to be used as inputs to a modified SWM. Rather than stacking 1500 MHz into a single signal, as is done with signals  202 - 208 , a larger number of outputs  312 - 326  are used. Although eight outputs  312 - 326  are shown, a larger or smaller number of outputs  312 - 326  are possible without departing from the scope of the present invention. 
     As shown in  FIG. 3B , LNB 1  signal  312 , LNB 2  signal  314 , LNB 5  signal  320 , and LNB 6  signal  322  each comprise two 500 MHz signals, each 500 MHz signal corresponding to a Ka-LO band signal and a Ka-HI band signal. This arrangement is appropriate if the A/D has enough front end bandwidth to subsample the Ka-Hi band in its 2 nd  Nyquist Band. Alternatively, separate independent IF conversion for the Ka-Lo and Ka-Hi can be provided for more traditional 1 st  Nyquist band sampling of each signal. LNB 3  signal  316 , LNB 4  signal  318 , LNB 7  signal  324 , and LNB 8  signal  326 , on the other hand, are 500 MHz signals, each corresponding to a Ku-band signal. To facilitate A/D conversion, the local oscillators  328  can be modified so that each of the signals  312 - 326  can have a desired and, possibly, different tunable starting frequency from 10 to 100 MHz, or beyond these limits if desired. Additional mixing can be added to achieve an offset frequency start for one or more of the signals  312 - 326  if desired. It is also possible within the scope of the present invention to implement tuning for signals  312 - 326  within the digital domain if desired. 
       FIG. 4  illustrates an embodiment of the Analog-to-Digital subsystem functionality of the present invention. 
       FIG. 4  shows the A/D subsystem  400  functions, where the sampling frequency Fs is adjustable, typically from 1.02 GHz to 1.2 GHz, but other frequencies and ranges are possible within the scope of the present invention. Each signal  312 - 326  is placed through a filter  402 , and then into an A/D converter  404 , which produce the output signals  406 - 428 . The typical maximal sampling rate of the A/D converters  404 , FsMAX, is typically 1.35 GHz, but other rates are possible without departing from the scope of the present invention. The Ka HI signals are typically sub-sampled, so the analog paths should have a useable bandwidth of FsMAX+500 MHz, which is typically 1.85 GHz. Alternatively if separate IF conversion is provided for the Ka-Hi signal then traditional 1 st  Nyquist band sampling and Lo-Pass anti-alias filters would be employed. 
       FIG. 5  illustrates a block diagram of the DSWM Channelizer in accordance with one or more embodiments of the present invention. 
     Each input “x”  500 ,  502 , etc., receives one of the A/D converter  404  outputs  406 - 428 , and there can be extra inputs x  500 ,  502 , etc., to allow for expansion of the system. Each typical input x  500  is typically channelized into uniformly spaced K filters  504 . The portion k  506  of the K filters  504  actually utilized may be different for different LNB paths, and, typically, k=12 for Ka HI, k=12 for Ka LO, and k=16 for Ku. L, the number of filter bands on the output side, is set to be equal to the number of stacked carriers desired at the stacked output  210 . 
     There are many embodiments within the scope of the present invention that can create computationally efficient DSP architectures. Another example embodiment of the present invention uses multi-rate poly phase techniques and takes advantage of the correspondence between the complex mixing and the Fast Fourier Transform (FFT), as shown in  FIG. 6 . 
       FIG. 6  illustrates a coarse granularity channelizer in accordance with one or more embodiments of the present invention. 
     In  FIG. 6 , system  600  shows inputs  406 - 428 , each entering a Real to Complex Hilbert transformer  600 . Other types of Real to Complex transforms of the signal inputs  406 - 428  are also possible within the scope of the present invention. The K filters  604  for each of the inputs  406 - 428  are set to 16, but other settings are possible within the scope of the present invention, including different values for K and x for each input  406 - 428 . If the transponder count and spacing are uniform and well known, then K can be set to the number of transponders, etc. If, however, the number of transponders varies from satellite to satellite and/or the spacing of these transponders is not uniform, then a number of filters  604  greater than the maximum number of transponders to be encountered is typically used to allow for expansion. Further, the filters are then expanded or contracted by slight adjustments in the sampling frequency and by slight shifts in the down-converter LO frequency used in the LNB. By adjustment of both the down-converter LO of the LNB and sampling rate, any channel spacings can be accommodated in any of the LNB outputs. 
     The signals  406 - 428 , after being filtered by filters  602  are subjected to Fast Fourier Transforms (FFT) in FFTs  606 , and then selected, reordered, and combined in multiplexer  608 . The combined signal is further processed to generate stacked output  210 . Alternatively, the present invention can use a non-maximally decimated filter bank of overlapping filters. This approach, along with the added technique of near-perfect reconstruction techniques simplify the fine granularity design. 
     A choice of L=32 and K=16 are illustrative only. The choice of the power of 2 composite K and L is chosen to simplify the FFT hardware implementation. Other approaches or arrangements may also be used within the scope of the present invention. 
     ADC ENOB Considerations 
     Consider any one of the 500 MHz LNB signals  406 - 428 . Since each signal is a composite of many signals, each signal is approximately Gaussian distributed. Given this approximation, the attack point on the ADC can be set low enough so that probability of the input exceeding the full scale deflection is controlled and thereby minimize the clipping noise effects. Similarly the attack point can be set high enough so that the ADC is driven hard enough to minimize the effects of quantization noise. The balancing of these two noise affects yields an optimal attack point (sometimes referred to as backoff point). For any given backoff point then the ADC Effective Number of Bits (ENOB) will determine the Signal to Quantization Noise Floor. For anticipated transponder loadings an ENOB of 8 bits is more than sufficient. Since A/D are available at approximately this ENOB while operating well in excess of 1 Gsps, the current state of the art in ADCs is sufficient to implement the designs described by this invention. However, an accounting of the Noise-Power Ratio of the finite word processing and additional details of the noise bandwidth and system settings may allow the ENOB to be relaxed from a 8 bit setting and still be within the scope of the present invention. 
     Fine Granularity Design 
       FIG. 7  illustrates a fine granularity channelizer implementation in accordance with one or more embodiments of the present invention. 
     A fine granularity design using perfect or near perfect reconstruction polyphase filtering techniques has advantages over the coarse granularity approach. For such an approach, the spectrum of each of the incoming 500 MHz blocks is subdivided much more finely than in the coarse granularity design. Where in the coarse granularity design the goal is to create a number of filters greater than or equal to the maximum number of expected transponders, in the fine granularity design the goal is to divide up the spectrum into smaller pieces. This can be done in such a way that the fine pieces of spectrum can be “glued” back together to yield a nearly perfect reconstruction of any arbitrary spectral bandwidth within the granularity specified for the design. Employment of non-maximal decimation techniques can be used to simplify filter design if desired. An illustrative design for a fine granularity system is shown in  FIG. 7 . 
     The system  700  shown in  FIG. 7  has four output signals  702 ,  704 ,  706 , and  708 . Outputs  702  and  704  are typically used to create two 500 MHz blocks of single wire bandwidth. Outputs  702  and  704  can then be power combined onto a single wire interface and thus replicate the output of related SWM designs, except that they provide more channels within the same physical bandwidth. 
     The other outputs  706  and  708  depicts two illustrative embodiments that include shared demod assets that do not expend any of the Single Wire Bandwidth. Other embodiments are possible within the scope of the present invention. Optionally, only one output  706  or  708  can be implemented, or other embodiments can be implemented alone or in any combination, without departing from the scope of the present invention. Output  706  illustrates an approach having an additional internal single wire interface which drives conventional receiver/demod inputs. 
     Output  708  illustrates an approach where the receiver portion of the receiver demod chips can be eliminated at the same time as the last up-conversion of the processed signals. 
     These chips correspond to I/Q near baseband demodulation. Other configurations are possible if some of the output polyphase filtering is incorporated directly on the demod chip for individual true baseband I/Q processing. In both outputs  706  and  708 , the output of the shared demod resources is Satellite Communicator Identification Code (SCID) filtered data which is networked onto any suitable physical layer and/or network layer protocols for distribution throughout the house or Multiple Dwelling Unit (MDU). The outputs  706  and  708  are Internet Protocol (IP) type outputs, or similar, that can be output over ethernet cabling, local area networks, RF, or other similar interfaces as desired, without departing from the scope of the present invention. 
     DRO Frequency Drift Control 
       FIG. 8  illustrates a functional block diagram of processing of one 500 MHz band in accordance with one or more embodiments of the present invention. 
     Advances in Digital Signal Processing (DSP) techniques, specifically, the speed at which DSP techniques can now take place, allow for more efficient bandwidth packing of signals on a single wire interface between ODU  108  and receiver  112 . However, in order to take advantage of this tighter signal packing, the frequency stability of frequency sources, in this case, the frequency source used to initially down-convert signals  120  into an intermediate frequency must be similarly improved. The present invention allows for estimation of frequency drift of the frequency source used for down-conversion, as well as allowing for the ability to measure frequency domain errors which will allow for diagnostic feedback on the state of system  100 . 
     As shown in  FIG. 8 , signals  120  are received at ODU  108 , and then are passed on to receiver  112 . Within ODU  108 , an antenna  800  receives the signals  120  at both Ku-band and Ka-band (e.g., approximately 20 GHz), which are mixed at mixer  802  with the output of DRO  804 . This mixing process results in an output of mixer  802  of the sum and difference frequencies of signals  120  and DRO  804 , and the frequency of DRO  804  is chosen to down-convert signals  120  to a frequency convenient for A/D conversion. 
     These signals are then typically filtered through a Band Pass Filter (BPF)  808  to remove harmonics and other unwanted signals, and then typically passed to Automatic Gain Control (AGC)  810  circuitry to normalize the signals  806  to a common signal strength. Such a common strength is typically desirable to properly excite Analog-to-Digital (A/D) converter  812  to set the attack point as previously described in the ENOB Considerations discussion. 
     A/D  812  is a high-speed A/D converter, in that it can run at 1 Gigasample per second (GSPS) or higher. Such speed is needed because the typical bandwidth of the signals  806  is 500 MHz, and to sample at the minimum Nyquist rate for such signals requires a minimum of twice the bandwidth, or 1 GSPS. A/D  812 , thanks to recent advances in A/D design, can now achieve 1 GSPS rates, and higher. Once digitized by A/D  812 , Digital Signal Processor (DSP)  814  processes the signals and converts them back to analog signals in the Digital-to-Analog Converter (DAC)  816 . Such signals are then mixed at mixer  818  with a local oscillator  820  and forwarded on to receiver  112 . Optionally, this last mixer can be avoided if a DAC with sufficient BW is employed such that it can directly cover the desired single wire interface band. 
     Within DSP  814 , the signals can be sent to a drift estimator  822 , where drift of the frequency of signals  806  can be sensed. Sensing drifts from various sources are achievable within the scope of the present invention, e.g., drifts which affect single carriers within a given bandwidth of signals or drifts which affect an aggregate of carriers, etc., however, the primary cause of drift of signals  806  is the DRO  804  frequency drift with temperature and age. DRO  804  is located at the dish antenna  800  of ODU  108 , and as the sun heats the dish antenna  800 , DRO  804  warms up which causes drift, and as the sun goes down dish antenna  800  cools down, cooling down DRO  804 , again causing drift in frequency. Further, aging of the DRO  804  will also cause a frequency drift in the DRO  804  output. 
     The drift estimator  822  can be of several varieties. One solution that can be used for drift estimator  822  is to create a discriminator based on a matched filter design, such that the expected type and distribution of transponders within the aggregate bandwidth of signals  806  is sensed. Such a discriminator can be developed for individual transponders if desired, or the edges of the bandwidth can be sensed. For transponders with symmetrical properties, the discriminator can be a matched filter with a sign reversal about the center frequency, however, if the purpose is to drive the average drift of the DRO  804  to zero, a matched filter for all of the transponders within the aggregate bandwidth can be utilized. Although the drift estimator algorithm may result in a biased estimate of each transponder&#39;s frequency that is down-converted by the DRO  804 , the DRO  804  is tuned by the aggregate of the transponder drift discriminants (biased or not) and will therefore tend to average out the biasing effects due to any individual transponder gain slopes and ripples. 
     The present invention also allows for system  100 , via ODU  108  (also called a Single-Wire Multiswitch ODU or SWM-ODU) to recognize large frequency deviations and frequency trends over the life of ODU  108 . As DRO  804  frequencies drift over the life of DRO  804 , the present invention can provide not only feedback to DRO  804 , but to system providers via receiver  112  callbacks or other communications between receiver  112  and system providers. Receivers  112  typically comprise a high-speed internet or other interface to allow communications between the system provider and individual receivers  112 ; reporting system  100  health issues can now, through the use of the present invention, be undertaken by processors in receiver  112  and/or DSP  814 . 
     The outputs of drift estimator  822  are then converted to analog voltages in DAC  824 , and then fed back to DRO  804  via a control input  826  to DRO  804 . In essence, DRO  804  acts like a Voltage Controlled Oscillator (VCO) at this point, where the voltage applied at  826  controls the frequency output of DRO  804 . 
     Although the use of DSP outputs to sense and correct frequency drift faults is emphasized herein, other uses of DSP  814  outputs to recognize other faults and to take other corrective actions are possible within the scope of the present invention. In particular, by virtue of the architecture of the described processing key diagnostic information is available for other diagnostics. A mixed signal ASIC will typically include the AGCs  810 , ADCs  812 , DSP  814 , DACs  816 , and a general purpose processor implementing the Drift Estimator  822  matched filter discriminant. Thus the general purpose processor has access to information regarding the state of each of these subsystems. Beyond determining the frequency drift, the matched filter results coupled with knowledge of the AGC states provides important diagnostic information if no match is found over the possible range of drift frequencies of a given expected transponder. The system can thereby recognize the absence of expected transponders and can report this fault. This fault represents a fault in system  100  to deliver the missing transponder to the ODU  108 . The failure of the match filter to find a match for any transponder in a given 500 MHz band indicates the potential failure of one of the 500 MHz signal paths  312 ,  314 ,  318 ,  320 ,  322 ,  324 , and  326  or subsequent processing these signal paths indicating a potential hardware failure in the ODU  108 . In particular, if it is known that other ODUs  108  at other sites (homes) in the near vicinity of a given ODU  108  are not reporting faults, then it can be concluded by system  100  that the fault is isolated to the given ODU (“lone ODU”). In contrast, if all ODUs  108  in the vicinity of a given ODU  108  all report faults on many 500 MHz bands over a given short period of time followed by a complete recovery, the likely cause of the event is a weather generated event. In particular, if a collocated group of ODUs report faults occurring primarily or for longer periods of time on the Ka bands as opposed to the Ku bands then a weather event is very likely since weather affects Ka signals much more severely then the Ku signals. If a lone ODU  108  reports failures on many of its signal paths for extended periods of time then ODU  108  mis-pointing is indicated. During installation, the diagnostics described also have value. Analyzing a combination of matched filter and AGC results, information to aid in the initial ODU antenna pointing and verification of the final installation can be provided. The above examples, e.g., the use of state information from at least one of the AGC, the ADC, the DSP, and the DAC providing a diagnostic output for the system, where the diagnostic output comprises at least one of fault recognition, fault reporting, performance monitoring, installation aiding, and installation verification for the system  100 , are provided as examples of the class of diagnostics and installation verification aids possible by analysis of data related to the state of the subsystems of the given architecture. Any such diagnostics or installation aids derived from such state information is within the scope of this invention. 
     Although discussed with respect to voltage control of DRO  804 , other methods of control of signals  806 , and the effects of DRO  804  drift, can be accomplished with the present invention. For example, and not by way of limitation, a digital compensation within DSP  814  of the frequency offset, once estimated by drift estimator  822 , can also be achieved via feedback between drift estimator  822  and DSP  814 . 
     CONCLUSION 
     The present invention discloses systems and devices for controlling frequency drift in satellite broadcast systems. A receiver antenna system for a direct broadcast satellite signal communications system in accordance with one or more embodiments of the present invention comprises an oscillator, a mixer, coupled to the oscillator, for converting satellite signals at a first frequency to signals at an intermediate frequency, an analog-to-digital (A/D) converter, coupled to the mixer, for receiving the signals at the intermediate frequency and for converting the signals at the intermediate frequency at near-real-time to a digital data stream, a Digital Signal Processor (DSP), coupled to the A/D converter, for processing the digital data stream, and a drift estimator, coupled to the DSP, the drift estimator determining a frequency drift of the oscillator, wherein the receiver antenna system corrects the frequency drift of the oscillator using the determined frequency drift. 
     Such a system further optionally comprises the drift estimator driving a digital mixer within the DSP to compensate for the determined frequency drift, an output of the drift estimator being fed back to the oscillator to control the frequency drift of the oscillator, an automatic gain control coupled between the mixer and the A/D converter, the A/D converter sampling the signals at the intermediate frequency at a speed greater than 1 gigasample per second, the satellite signals being transmitted in at least the Ku-band of frequencies, the satellite signals being further transmitted in at least the Ka-band of frequencies, a Digital-to Analog Converter (DAC), coupled to the DSP, and an output of the DAC being mixed with a second oscillator, such that an output of the receiver antenna system is set to a desired band on a single wire interface. 
     A system for distributing a plurality of satellite signals on a single interface in accordance with one or more embodiments of the present invention comprises an oscillator for down-converting the plurality of satellite signals to signals at an intermediate frequency, an Automatic Gain Controller (AGC) for gain controlling the signals at the intermediate frequency, an analog-to-digital (A/D) converter, coupled to the AGC, for receiving the signals at the intermediate frequency, wherein the A/D converter directly samples the signals at the intermediate frequency, and a Digital Signal Processor (DSP), coupled to the A/D converter, wherein a first output of the DSP is used to determine the intermediate frequency and a second output of the DSP is an input to the single interface. 
     Such a system further optionally comprises a Digital-to-Analog Converter (DAC), coupled to the second output of the DSP, the first output of the DSP determining the intermediate frequency by controlling a frequency of the oscillator, the first output of the DSP driving a compensatory frequency shift by controlling a digital mixer internal to the DSP, the oscillator being a Dielectric Resonance Oscillator (DRO), the plurality of satellite signals being transmitted in a plurality of frequency bands, the plurality of frequency bands comprising a Ka-band and a Ku-band, the DRO down-converting the Ka-band to at least a first intermediate frequency band and the DRO down-converting the Ku-band to a second intermediate frequency band as convenient for A/D conversion, the A/D converter sampling the signals at the intermediate frequency at a rate greater than the Nyquist rate for the signals at the intermediate frequency, state information from at least one of the AGC, the ADC, the DSP, and the DAC providing a diagnostic output for the system, and the diagnostic output comprising at least one of fault recognition, fault reporting, performance monitoring, installation aiding, and installation verification for the system. 
     The foregoing description of the preferred embodiment of the invention has been presented for the purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise form disclosed. Many modifications and variations are possible in light of the above teaching. It is intended that the scope of the invention be limited not by this detailed description, but by the claims appended hereto and the full range of equivalents of the claims appended hereto.