Abstract:
An electric motor ( 40 ) has a permanent-magnet rotor ( 46 ) and an apparatus for generating a three-phase sinusoidal current (i 202 , i 204 , i 206 ) for supplying current to said motor ( 40 ), also a microprocessor ( 95 ) for executing the following steps: while the motor ( 40 ) is running at a substantially constant load, the motor is operated firstly at a predetermined operating voltage (U), and an amplitude of a current flowing to the motor is iteratively sampled, stored, and compared as applied voltage is decreased. If it is found, in this context, that the current flowing to the motor has not decreased as a result of reduction in the voltage amplitude, the motor ( 40 ) is operated at that current. If, however, it is found that the current flowing to the motor has decreased as a result of the reduction in the voltage delivered to the motor ( 40 ), the measurements and the comparison are repeated, optionally multiple times, in order to identify values for optimized efficiency.

Description:
CROSS-REFERENCES 
     This application is a section 371 of PCT/EP11/00354, filed Jan. 27, 2011 published Aug. 4, 2011 as WO 2011-092011-A, and further claims priority from German application DE 10 2010 006 337.1, the entire content of which is hereby incorporated by reference. 
     FIELD OF THE INVENTION 
     The invention relates to a method for improving efficiency in a multiphase motor, and it relates to a motor for carrying out such a method. The motor is preferably a three-phase permanent magnet synchronous motor (PMSM). 
     BACKGROUND 
     Good efficiency with little hardware complexity is desirable for the operation of an electronically commutated motor. 
     The efficiency of a motor is defined by
 
Efficiency= P   out   /P   in   (1)
 
     When the efficiency is at its maximum, the quotient P out /P in  must therefore also be at its maximum. 
     In the above equation,
 
 P   in   =U*I =electrical power absorbed by the motor  (2)
 
 P   out   =T*n =mechanical shaft power output  (3),
 
     where
     U=voltage   I=current   T=torque   n=rotation speed.   

     At a constant load torque T=constant and constant rotation speed n=constant, i.e. in a state of constant load that exists, for example, in the case of a fan in continuous operation, the variable component in equation (1) is the absorbed power level P in =U*I. 
     The voltage Û is normally constant, and the current I is thus the variable to be controlled. 
     The definition of the so-called air-gap torque is
 
 T   Mi ( t )= C   M *Ψ( t )* I ( t )  (4)
 
where
     T Mi =internal torque or air-gap torque of the motor   C M =machine constant   ψ=concatenated flux   I=current to the stator, e.g. current in one phase, or total current to the stator, as explained below.   

     The requirement that the curve for T Mi  be uniform, or “smooth,” yields the requirement that both the concatenated flux ψ and the current I should be sinusoidal. This results in the requirement that the phase relationship between current I and flux ψ be determined so that a maximum torque T Mi  is obtained. 
     If the stator flux ψ and stator current I in a three-phase synchronous motor are parallel vectors, the torque T generated by the motor is then equal to zero. If, on the other hand, the space vector is at right angles to the stator current, a maximum torque is then produced. This is similar to the situation with a direct-current motor. 
     Generating this right angle by a control procedure requires a control loop with feedback to the machine, indicating the position of the rotor. This feedback has often been implemented in synchronous machines using three Hall sensors. Today, in most cases, encoders (resolvers), optical incremental and absolute value sensors, or inductive sensors are used. Sensorless control systems can be carried out, in a context of block commutation, by measuring the back-EMF induced in the motor. 
     It is known, from the prior art, to operate a three-phase synchronous motor with good efficiency using field-oriented control (FOC). As depicted in  FIG. 8 , in this case the rotor position, and thus the phase relationship of the flux, is ascertained either via a rotor position sensor or using sensorless methods, e.g. a so-called “observer” design. 
     In field-oriented control (FOC), the measured phase currents are broken down by matrix operations (Park-Clarke transformation or inverse Park-Clarke transformation) into two components: a field-forming part id and a torque-forming part iq. This type of subdivision into components makes it possible in FOC to modify or control the field-forming variable id independently of the torque-forming variable iq. The field-forming variable is equal to zero at the point of maximum efficiency. This results in a special case that can easily be implemented with no need to carry out complex matrix operations, i.e. FOC can be dispensed with, in this special case. 
     Because matrix operations are not necessary, a simple microprocessor can be used, whereas expensive microprocessors having a digital signal processor (DSP) would otherwise be required for FOC. 
     In this case, a brief measurement operation can be used to determine the phase relationship between flux ψ and motor current I which results in the maximum torque T. The equation is: 
     
       
         
           
             
               F 
               ⁡ 
               
                 ( 
                 x 
                 ) 
               
             
             = 
             
               
                 
                   ∫ 
                   0 
                   π 
                 
                 ⁢ 
                 
                   
                     sin 
                     ⁡ 
                     
                       ( 
                       x 
                       ) 
                     
                   
                   * 
                   
                     sin 
                     ⁡ 
                     
                       ( 
                       
                         x 
                         + 
                         α 
                       
                       ) 
                     
                   
                 
               
               = 
               
                 
                   
                     
                       cos 
                       ⁡ 
                       
                         ( 
                         α 
                         ) 
                       
                     
                     * 
                     π 
                   
                   2 
                 
                 = 
                 
                   max 
                   . 
                 
               
             
           
         
       
     
     where:
     x=rotation angle of rotor, usually measured in radians   α=phase difference between current I and flux ψ (see  FIG. 1 )   

     1. Necessary: 
     
       
         
           
             
               
                 f 
                 ′ 
               
               ⁡ 
               
                 ( 
                 α 
                 ) 
               
             
             = 
             
               
                 
                   ⅆ 
                   
                     ⅆ 
                     
                       ( 
                       α 
                       ) 
                     
                   
                 
                 ⁢ 
                 
                   
                     
                       cos 
                       ⁡ 
                       
                         ( 
                         α 
                         ) 
                       
                     
                     * 
                     π 
                   
                   2 
                 
               
               = 
               
                 
                   0 
                   ⇒ 
                   
                     
                       
                         - 
                         
                           sin 
                           ⁡ 
                           
                             ( 
                             α 
                             ) 
                           
                         
                       
                       * 
                       π 
                     
                     2 
                   
                 
                 = 
                 
                   
                     0 
                     ⇒ 
                     α 
                   
                   = 
                   0 
                 
               
             
           
         
       
     
     2. Sufficient: 
     
       
         
           
             
               
                 
                   f 
                   ′′ 
                 
                 ⁡ 
                 
                   ( 
                   α 
                   ) 
                 
               
               = 
               
                 
                   
                     ⅆ 
                     
                       ⅆ 
                       
                         ( 
                         α 
                         ) 
                       
                     
                   
                   ⁢ 
                   
                     
                       
                         - 
                         
                           sin 
                           ⁡ 
                           
                             ( 
                             α 
                             ) 
                           
                         
                       
                       * 
                       π 
                     
                     2 
                   
                 
                 &gt; 
                 0 
               
             
             ⁢ 
             
                 
             
           
         
       
       
         
           
             for 
             ⁢ 
             
                 
             
           
         
       
       
         
           
             α 
             = 
             
               
                 0 
                 ⇒ 
                 
                   
                     f 
                     ′′ 
                   
                   ⁡ 
                   
                     ( 
                     0 
                     ) 
                   
                 
               
               = 
               
                 
                   
                     
                       cos 
                       ⁡ 
                       
                         ( 
                         0 
                         ) 
                       
                     
                     * 
                     π 
                   
                   2 
                 
                 = 
                 
                   π 
                   2 
                 
               
             
           
         
       
     
     This calculation yields the requirement for a sinusoidal current, which must be in-phase with the concatenated flux ψ, in order for efficiency to become optimal. 
     This is illustrated in  FIGS. 1 and 2 . The number  20  illustrates the overlap between a phase current, e.g. i_U, and the variable ψ. It is evident that the area  20  reaches its maximum when ψ and i_U are in-phase. 
     If the flux ψ and current I deviate from the sinusoidal shape,  FIG. 1  likewise indicates the need for the in-phase condition, in order to obtain improved efficiency. 
     SUMMARY OF THE INVENTION 
     It is therefore an object of the invention to make available a novel method of operating a PMSM (Permanent Magnet Synchronous Motor), and a novel PMSM which operates according to the method. 
     According to the invention, this object is achieved by a method wherein a control circuit applies three phases of sinusoidal current to the motor windings, and the motor is operated at constant load, while a microprocessor reduces the applied motor voltage until motor current sampling reveals that an optimal efficiency level has been achieved, the optimizing steps being repeated if a change in motor load is detected. 
    
    
     
       BRIEF FIGURE DESCRIPTION 
       Further details and advantageous refinements of the invention are evident from the exemplifying embodiments, in no way to be understood as a limitation of the invention, that are described below and depicted in the drawings. 
         FIG. 1  shows the phase relationships between the concatenated flux ψ (corresponding approximately to motor voltage U) and the current i_U in the stator phase U of a PMSM, a phase shift α being present which impedes optimum efficiency; 
         FIG. 2  shows a phase relationship with optimized efficiency; the flux ψ and phase current i_U are in this case in phase, and the efficiency is in the vicinity of its optimum; 
         FIG. 3  shows the construction of a circuit for automatically optimizing the efficiency of a PMSM; 
         FIG. 4  depicts motor  40  with a delta circuit configuration of stator winding  44 ; 
         FIG. 5  shows an algorithm for ascertaining electrical values in order to achieve optimum efficiency in the PMSM; 
         FIG. 6  shows the schematic configuration of a three-phase motor with permanent-magnet excitation; 
         FIG. 7  is a diagram explaining the generation of a three-phase system; 
         FIG. 8  depicts the digitization of a sinusoidal voltage; and 
         FIG. 9  depicts an arrangement according to the prior art. 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 3  shows an exemplifying embodiment of a multiphase permanent-magnet synchronous motor  40  that, during operation, is supplied with a multiphase alternating voltage via an inverter  42 . A three-phase motor  40  having a star-configured stator winding  44  is depicted. A delta configuration is alternatively possible (see  FIG. 4 ). Other numbers of phases are likewise possible. Motor  40  has a symbolically depicted permanent-magnet rotor  46  that is depicted as having two poles but can of course have four, six, eight, ten, etc. poles. This rotor  46  preferably has a sinusoidal magnetization, since motor  40  then produces a largely constant torque when sinusoidal stator currents are used. 
       FIG. 1  and  FIG. 2  each depict the strand current i_U, and the latter is therefore also shown in  FIG. 3 . Motor  40  can have any configuration, e.g. internal-rotor motor, external-rotor motor, motor with planar air gap, etc. 
     Motor  40  serves, for example, to drive a fan  48 , which represents a largely constant load during operation and serves, for example, to cool an electronic device, e.g. a computer. Another application is, for example, driving a pump for liquid cooling of a processor, in which case the load is likewise largely constant. 
     In the exemplifying embodiment, a DC link circuit  50  is used. This can be connected, for example, to the exchange battery of a telephone exchange. As depicted, link circuit  50  is powered, via a rectifier  52 , from an alternating-current grid  54 . 
     The current I in link circuit  50  is measured or sampled at a measurement element  56 , e.g. a measuring resistor or a current transformer (see  FIG. 3 ). 
     An FOC (Field Oriented Control) component  42  receives, at its input  60 , a (variable) voltage U. The amplitude Û of this voltage is modified, in steps, during efficiency optimization. Component  42  furthermore receives, at its input  62 , an angle β that defines the speed of the rotating field generated in motor  40 . Because motor  40  is synchronous, it does not inherently require a rotor position sensor, but such a sensor may be necessary, in order to allow continuous monitoring as to whether rotor  46  is rotating during operation, or whether it has come to a stop because its pull-out torque has been exceeded. The FOC component  42  controls a three-phase inverter  43 , to which winding  44  of motor  40  is connected. 
     Motor  40  has a rotation speed controller  70  for specifying the frequency of the rotating field to be generated by components  42 ,  43 . The output signal of controller  70  is applied to a summing unit  72  that produces a rotation speed ramp from zero speed to a speed n, i.e. a slow rise in rotation speed. The output signal of summing unit  72  is applied to an integrator  74 , which generates the ramp function and whose output signal is applied to a negative input of summing unit  72 . 
     The motor has a switch  78  having two switch positions: “1” and “2”. 
     The switch position “1” is used
     a) when the motor is starting up, i.e. upon acceleration, and   b) when changes in rotation speed occur.   

     This switch position “1” is also referred to as “adjusted” operation, since the motor is adjusted to specific operating data. 
     The switch position “2” signifies a seeking function, and is set when efficiency needs to be optimized by a seeking function (see  FIGS. 1 and 2 ). 
     A signal for the rotation speed n is obtained at the output of integrator  74 , and is applied to an element  76  that generates a rotation speed-dependent factor P. This determines the voltage amplitude according to the formula
 
 Û=n*P   (5),
 
i.e. the voltage amplitude Û increases as the rotation speed rises. In “adjusted” mode, this amplitude is applied via switch  78  (switch position 1) to input  60  of inverter  42 , with the result that the latter operates at an optimum working point.
 
     Position “2” of switch  78  is set when the efficiency is to be optimized by a seeking function. In this position, terminal  60  is connected to a summing element  80  which serves to calculate the amplitude Û when the seeking method for the optimum efficiency is activated. 
     The signal n*P (equation 5) is applied to a positive input of summing element  80 . A “zero” signal is applied to a negative input via a switch  82  when switch  78  is in position “1”. In position “2” (seek mode), a “1” signal is applied to this input. The output signal of an integrator  84  is applied to another negative input of summing element  80 . This causes a reduction in amplitude in continuous operation. 
     The output signal of a multiplier  86  is applied to the input of integrator  84 . This signal serves to generate an amplitude reduction ramp, which generates a value of
     0*rotation speed-dependent reduction, or   1*rotation speed-dependent reduction.   

     A rotation speed-dependent factor P for the reduction of the amplitude Û is applied to the one input of multiplier  86 , from a transducer  88 , to whose input the rotation speed n is applied. 
     The output signal of a decision element  90  is applied to the other input of multiplier  86 , said element deciding between
     the criterion “greater than” (&gt;) and   the criterion “less than or equal to” (≦).   

     The output signal of a difference element  92  is applied to the input of decision element  90 , said element  92  serving to establish the difference between two successive measurements of current I. In other words, in the course of the seeking function, the amplitude of current I changes, until it has reached a minimum, and the current I rises again, once that minimum is reached. 
     An integrator  94  integrates the rotation speed value n and generates, at its output, the angle value β that is applied to input  62  of inverter  42 . 
     The components of  FIG. 3  that are surrounded by a dot-dash line  95  are constituents of a microprocessor  95 . In the exemplifying embodiment, an eight-bit microcontroller, having three PWM (Pulse Width Modulation) generators to generate the three phase currents, and having three sine-wave transducers to generate the three sine-wave voltages of the three-phase system, was used. This is described in  FIG. 6 . A suitable type is, for example, the PIC16F1938 of Microchip Technology, Inc. of Chandler, Ariz., USA, which has 3 so-called “Capture/Compare/PWM” modules built-in. 
     Mode of Operation 
     When motor  40  starts, a rotating voltage field is generated by inverter  42 ,  43 . This field has an amplitude which is sufficiently high to generate a torque that is sufficient to start motor  40 . 
     When motor  40  is running, the voltage amplitude Û of the rotating field that is to be outputted is then lowered, in steps. The result is, firstly, to decrease the current amplitude, which reaches its minimum at the point of optimum efficiency. 
     The load on the motor should, in this context, be as constant as possible. 
     Once the optimum efficiency is reached, the voltage amplitude Û is held at the value that was reached, as long as no elevation in current (above a predetermined threshold) occurs. If such an elevation in current does occur, it is the consequence of a change in load, and a new operating point is then set, i.e. the above-described seeking function is repeated. 
     The present invention thus exploits the property according to which the amplitude of the current I becomes minimal at the point of optimum efficiency. In other words, this means that, in order to achieve the same operating state for a different position (angle α in  FIG. 1 ) of current I relative to flux (Psi), this operating state would be achievable only by a higher absolute value for the current amplitude, i.e. with a poorer efficiency. The current value used can be either a phase current, e.g. i_U, or the total current I flowing to motor  40 , or (in  FIG. 4 ) the current through resistor  56 , i.e. the current through semiconductor switch  52 . 
     Advantages that can be pointed out are, among others: 
     
         
         
           
             The method can be implemented with an inexpensive microprocessor  95 . 
             The method does not require any motor-specific parameters, and can thus be used with all motors of that type. (Sensorless methods, in contrast, require motor-specific parameters for flux determination.) 
             Environmental influences (temperature, humidity) and production tolerances have no influence. This contrasts with sensorless methods for determining rotor position, in which the motor-specific parameters can change as a result of the influence of temperature and because of production tolerances. (Such influences can result in error in estimating the flux and in specifying the controller target value.) 
           
         
       
    
       FIG. 5  schematically illustrates execution of an iteration S 100 , with which motor  40  is adjusted to an optimum working point for the instantaneous load. 
     At S 102 , a voltage amplitude Û=U START  is set in component  70  of  FIG. 3  (see  FIG. 1 ), and at S 104  this instantaneously active amplitude Û is stored, i.e. U active =Û, so that, at the next iteration, the voltage value Û used at that time can be stored again. 
     At S 106 , the current I new  that occurs at this voltage amplitude Û is measured and stored, i.e. I new =I. The measured current can be either the current in one strand of motor  40 , e.g. in  FIG. 3  the current i_U in strand U, or the total current I that is measured in  FIG. 3  at a measuring resistor  66 . With the latter variant, the losses are of course somewhat higher, i.e. a somewhat lower efficiency is obtained. The current in resistor  56  of  FIG. 4  is also suitable. 
     At S 108 , this current I new  is copied into a register I old , i.e. I old =I new , so that the next current value measured in the course of the iteration can then be stored in the register I new . 
     In S 110 , the voltage amplitude being used is reduced by a predetermined value U Delta , i.e.
 
 Û=Û   active   −U   Delta .  (6)
 
At S 112 , this new voltage amplitude Û is stored in the register for the value Û, i.e. Û active =Û.
 
     This reduced voltage amplitude results in a new value for the current I new , which either can be of the same magnitude as the previous current value I old , or can be smaller or larger than that value. This new current value I new  is measured in S 114  and stored in the register I new . 
     Step S 116  then checks whether I old  was greater than I new , i.e. whether the current has moved closer to the optimum value, or whether the values have remained the same, or whether I old  is less than I new , which would mean that the value is moving further away from the optimum. 
     If the response in S 116  is YES, then optimization is not yet complete, and the routine returns to S 108 , i.e. the current I new  measured in S 114  is copied into the register I old  and steps S 110 , S 112 , S 114 , and S 116  are repeated. 
     A state is ultimately reached in which I old  is no longer greater than I new , but instead is either of the same magnitude or is, in fact, smaller. In this case, the response in S 116  is NO, i.e. the optimum region (for the instantaneous load of motor  40 ) has been found, and the routine comes to an end at S 118 , because the optimum region has been ascertained. Motor  40  then runs at that voltage U active  from S 112  until, if applicable, the load changes. 
     In the event of load changes, the routine goes back to step S 102  and the entire iteration begins again, i.e. motor  40  then seeks, for the new load, a new optimized voltage Û at which the motor current (or the strand current) arrives at a minimum. 
       FIG. 6  shows, on the right, motor  40 , the stator of which has three phases  202 ,  204 ,  206 . Motor  40  has permanent-magnet rotor  46 , which is depicted as a four-pole rotor. Its poles are sinusoidally magnetized. One example of a rotor of this kind having sinusoidal magnetization is the rotor according to DE 100 20 946 A1, SCHNEIDER et al, published 15 Nov. 2001, assigned to Siemens AG. 
     The three phases  202 ,  204 ,  206  are supplied with three-phase current that is generated in the motor, the motor being automatically adjusted to good efficiency. 
     For this, μC 95 generates three sine-wave signals, namely:
 
sin  t  
 
sin( t+ 120°)
 
sin( t+ 240°).
 
     The frequency of these three signals is adjustable at μC 95 via a signal  250 . Because this frequency specifies the speed of the rotating field, and thus the rotation speed of rotor  40 , a rotation speed measurement is not necessary, except if separate rotation speed monitoring of motor  40  is desired, for example, in case it exceeds its pull-out torque and, as a result, comes to a halt. 
     The signals sin t, sin(t+120°), and sin(t+240°) are compared, in comparators  272 ,  274 ,  276 , with the triangular signal u  270  at the output of a triangular signal generator  268 , which is applied to the inverting inputs of the three comparators  272 ,  274 ,  276 . The associated sine-wave signal from μC 95 is applied, as shown in  FIG. 6 , to the respective non-inverting input of the associated comparator. The signals PWM 1 , PWM 2 , PWM 3  (that are shown in  FIG. 7 ) are then obtained, as the outputs of comparators  272 ,  274 ,  276 . 
     The signal PWM 1  is applied to a driver module  286  whose upper output  288  is connected to the gate of an n-channel MOSFET  290 , one terminal of which is connected to lead  50  at which the link circuit voltage U ZK  is present. Its other terminal is connected to strand  204 . 
     The lower output  294  of driver module  286  is connected to the gate of an n-channel MOSFET  296 , the upper terminal of which is likewise connected to strand  204 , and the lower terminal of which is connected via measuring resistor  264  to ground  300 . 
     The signal PWM 2  is delivered to a driver module  304 , the upper output  306  of which controls an upper n-channel MOSFET  308  and the lower output  310  of which controls a lower n-channel MOSFET  312 . The circuit corresponds to that of MOSFETs  290 ,  296 , but MOSFETs  308 ,  312  control strand  202 . 
     The signal PWM 3  is delivered to a driver module  316 , the upper output  318  of which controls an upper n-channel MOSFET  320  and the lower output  322  of which controls a lower n-channel MOSFET  324 . The circuit corresponds to that of MOSFETs  290 ,  296 , but MOSFETs  320 ,  324  control strand  206 . 
     When MOSFET  290  and MOSFET  324 , for example, are conductive simultaneously, a current flows from positive lead  50  through n-channel MOSFET  290 , strands  204 ,  206 , n-channel MOSFET  324 , and measuring resistor  264  to ground  300 . As shown, recovery diodes are connected in antiparallel with the MOSFETs. 
       FIG. 7   a ) shows the signal PWM 1  in a highly schematic manner.  FIG. 7   b ) shows the current i 204  through phase  204  that is caused by the signal PWM 1 . This is a sinusoidal current that is brought about by the totality of the many switchover operations that take place as rotor  46  rotates. 
       FIG. 7   c ) shows the signal PWM 2  at the output of comparator  274 , and  FIG. 7   d ) shows the current i 202  through strand  202 . This current is likewise sinusoidal, and is offset 120° in phase with respect to strand i 204 . 
       FIG. 7   e ) shows the signal PWM 3  at output  282  of comparator  276 , and  FIG. 7   f ) shows the current i 206  through strand  206 . This current is offset 240° in phase with respect to the current i 204 , and is likewise sinusoidal. 
     The three sinusoidal currents i 204 , i 202 , and i 206  together constitute a three-phase system, and generate a rotating field that drives the permanent-magnet rotor  46  at the rotation frequency of that rotating field, as already explained. Because the magnetization of rotor  46  is sinusoidal, a largely constant torque is produced, and that torque is achieved with little complexity. In particular, there is no need for complicated and expensive rotation angle sensors, and motor  40  operates at optimized efficiency. 
       FIG. 8   a ) shows, in schematic form, the signal u 270  that is generated by triangular signal generator  268 . The frequency of the signal u 270  is assumed here to be 20 kHz, i.e. one triangle of the triangular signal u 270  has a period length of 50 μs. The first triangle, labeled  338 , begins at the 0 μs time, reaches its maximum at 25 μs, and returns to zero at 50 μs. It is therefore symmetrical, and preferably has the shape of an isosceles triangle. Its frequency is also high in relation to the frequency of the signal sin t. 
     As long as the latter signal is greater than u 270 , the signal PWM 1  depicted in  FIG. 8   b ) is HIGH. If Hl becomes less than u 270 , PWM  1  then takes on the LOW value. This results in the profile (depicted in  FIG. 8   b )) that is typical for PWM 1 , where the pulse duty factor is high to the left and right, e.g. 90%, while in the middle it has a value of approximately 10%, producing a largely symmetrical profile in  FIG. 8   b ). 
     Be it noted that  FIG. 8  shows a simplification, since in reality over one hundred triangles of the signal u 270  are obtained for the duration of one period of the signal sin t, but this could not be depicted graphically. 
     The symmetrical triangular shape of the pulses of the signal u 270  yields the advantage that the PWM signals according to  FIG. 8   b ) are always located substantially symmetrically with respect to the maximum of a triangle. 
     The invention thus provides a simple configuration for a three-phase motor  40  of this kind, the rotation speed being specifiable by the signal at input  250  ( FIG. 6 ). The current for the optimization procedure according to  FIG. 4  is normally measured or sampled at only a single bridge arm, e.g. by means of measuring resistor  56  depicted there, so that good efficiency is obtained. 
     Numerous variants and modifications are of course possible, within the scope of the present invention.