Abstract:
A wideband antenna includes a plurality of radiating elements arranged in an array and a feed network. The feed network includes at least one frequency dependent power divider for varying the amplitude of a signal provided to at least two of the plurality of radiating elements as a function of a frequency of a signal. The feed network may further comprise a plurality of inputs and the antenna may produce a plurality of beams. The frequency dependent divider may comprise a power divider having a first output and a second output, a 90° hybrid, having a first input coupled to the first output of the power divider, and a second input, and a delay line, coupled between the second output of the power divider and the second input of the 90° hybrid.

Description:
RELATED APPLICATIONS 
       [0001]    This application claims priority to U.S. Provisional Patent Application No. 61/845,616, filed Jul. 12, 2013 and titled “Wideband Twin Beam Antenna Array”, the entire disclosure of which is incorporated by reference. 
     
    
     FIELD OF THE INVENTION 
       [0002]    The present invention generally relates to radio communication. More particularly, the invention relates to wideband multi-beam antennas for cellular communication systems. 
       BACKGROUND OF THE INVENTION 
       [0003]    For common cellular applications where a given site has three sectors, each serviced by one antenna, each antenna usually has 65 degree azimuth Half Power Beamwidth (HPBW). Six sector cells may be employed at such sites to increase system capacity. Antennas with 33 degree and 45 degree HPBW are the most common for 6 sector applications. However, replacing three antennas with six antennas (each of which is 2 times wider than a common 65 degree antenna) is not a compact and low cost solution. 
         [0004]    Dual-beam (or multi-beam antennas) may be used to reduce number of antennas on the tower. One key aspect of a multi-beam antenna is the beam forming network (BFN). An example of a known dual-beam antenna from prior art may be found in Encyclopedia for RF and Microwave engineering, 2005 John Wiley and Sons, pp. 335-339. In the known dual beam antenna, a two beam by two column (2×2) BFN is used. The main drawback of this prior art antenna is high levels of side and backlobes (about −10 dB), which is not acceptable for modern systems due high interference. Another drawback is poor coverage: more than 50% of radiated power is wasted out of the desired two 60 degree sectors. 
         [0005]    Another dual-beam prior art solution may be found in U.S. Pat. No. 8,237,619. This example includes a three column array. However, it also has very high sidelobes (about −9 dB) and relatively narrow operational band (1.7-2.1 GHz). In modem telecommunication systems it is desirable to include the LTE 2.6 GHz band, and a cover a bandwidth of 1.69-2.69 GHz, which is about 2.5 times wider bandwidth than in known solutions. 
         [0006]    Another dual-beam prior art solution is U.S. Patent Pub. No. 2011/0205119, which is incorporated by reference. In the &#39;119 application, improved azimuth suppression and wideband operation is achieved by including different antenna sub-arrays into an antenna array. For example, the antenna may include a two beam by three column array and a two beam by two column array. However, bandwidth is also limited for this prior art solution. 
         [0007]    Consequently, there is a need to provide an improved antenna, with more consistent HPBW across a wider frequency bandwidth, better azimuth sidelobe suppression, with better coverage of desired sector and less interference with other sectors. 
       SUMMARY OF THE INVENTION 
       [0008]    In one example, a Frequency Dependent Divider (FDD) is included in the beam forming network for wideband beam forming and creation of frequency reconfigurable antennas. The FDD, integrated in the beam forming network, provides for changes in amplitude distribution on array elements with respect to frequency. 
         [0009]    The FDD may be configured to change the power ratio on its outputs with respect to frequency. For example, for lowest frequency, all power goes to port  1 , for highest frequency all power goes to port  2 , and for central frequency power is about equal for both ports of FDD. In one aspect of the invention, a new compact frequency-dependent divider is proposed, comprising 3 dB divider, 90 degree hybrid and delay line between them. But other schemes of FDD (for example, filters) may be used. 
         [0010]    In another aspect of invention, a combination of  4  element and  5  element linear arrays is used to create a dual beam antenna with stabilized HPBW (+/−2 degree from nominal in 1.7 to 2.7 GHz), stabilized beam position (+/−1 degree from nominal) with low sidelobe level in azimuth an elevation plane for both beams. 
         [0011]    According to one aspect of the invention, a wideband antenna includes a plurality of radiating elements arranged in an array and a feed network. The feed network has at least one input and a plurality of outputs coupled to the plurality of radiating elements. The feed network further includes at least one frequency dependent power divider for varying the amplitude of a signal provided to at least two of the plurality of radiating elements as a function of a frequency of a signal. In one example, the feed network increasingly tapers a power distribution to radiating elements at each end of the wideband array as a function of increasing frequency. Other power distribution schemes may also be advantageous and are described in the detailed description and drawings. 
         [0012]    In another example, the feed network further comprises a plurality of inputs and the antenna produces a plurality of beams. For example, the feed network may further include a 90° hybrid having two inputs and the antenna produces two beams. 
         [0013]    The frequency dependent divider may comprise a power divider having a first output and a second output, a 90° hybrid, having a first input coupled to the first output of the power divider, and a second input, and a delay line, coupled between the second output of the power divider and the second input of the 90° hybrid. The delay line may be a regular transmission line, a regular transmission line combined with a Shiffman phase shifter, a transmission line incorporating a series inductances and parallel capacitors, or other suitable structure. 
         [0014]    In one example, a dual beam wideband array is provided. The array includes at least first, second and third radiating elements, the first, second and third radiating elements. A 90° hybrid having a first beam input and a second beam input is included. The 90° hybrid has a first output and a second output, the second output being coupled to the first radiating element. The array further includes a first frequency dependent power divider having an input coupled to the first output of the first 90° hybrid and a first output coupled to the second radiating element and a second output coupled to the third radiating element. The second output of the frequency dependent power divider is coupled to the third radiating element. 
         [0015]    In one example, the dual beam wideband array consists of three radiating elements, and the second and third radiating elements are at opposite ends of the array. In another example, the dual beam wideband array consists of four radiating elements, and the second and third radiating elements are at one end of the array. In another example, The dual beam wideband array further includes a second frequency-dependent divider and a power divider coupling the first output of the 90° hybrid to the respective inputs of the first and second frequency dependent dividers, and the array further comprises a fourth radiating element and a fifth radiating element. The second frequency dependent divider is coupled to fourth and fifth radiating elements. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0016]      FIG. 1A  is schematic diagram of a wideband beam forming network and array according to one aspect of the present invention. 
           [0017]      FIG. 1B  illustrates a simulated radiation pattern of one beam produced by the beam forming network and array of  FIG. 1A . 
           [0018]      FIG. 2A  is schematic diagram of another example of a beam forming network and array according to another aspect of the present invention. 
           [0019]      FIG. 2B  illustrates a simulated radiation pattern one beam produced by the network illustrated in  FIG. 2A . 
           [0020]      FIG. 3A  is schematic diagram of another example of a beam forming network and array according to another aspect of the present invention. 
           [0021]      FIG. 3B  illustrates a simulated radiation pattern of one beam produced by the beam forming network and array illustrated in  FIG. 3A . 
           [0022]      FIG. 3C  illustrates simulated radiation patterns for two beams produced by the beam forming network and array illustrated in  FIG. 3A . 
           [0023]      FIG. 4A  illustrates a first example of a delay line according to one aspect of the present invention. 
           [0024]      FIG. 4B  illustrates a second example of a delay line according to one aspect of the present invention. 
           [0025]      FIG. 4C  illustrates a third example of a delay line according to one aspect of the present invention. 
           [0026]      FIG. 5A  illustrates a first example of how beam forming networks according to the present invention and having different numbers of radiating elements may be combined in an antenna array. 
           [0027]      FIG. 5B  illustrates a second example of how beam forming networks according to the present invention and having different numbers of radiating elements may be combined in an antenna array. 
           [0028]      FIG. 5C  illustrates a third example of how beam forming networks according to the present invention and having different numbers of radiating elements may be combined in an antenna array. 
           [0029]      FIG. 5D  illustrates a fourth example of how beam forming networks according to the present invention and having different numbers of radiating elements may be combined in an antenna array. 
       
    
    
     DETAILED DESCRIPTION 
       [0030]    One aspect of the present invention is to compensate for the array factor of a phased-array multi-beam antenna with changing frequency. For example,  FIG. 1A  shows schematic diagram of beam forming network  10  and an array  50  of three radiating elements  51 ,  52  and  53 . This example accepts two inputs, Beam  1  and Beam  2 , and produces two beams. The beam forming network  10  comprises a 90° hybrid  12 , a frequency dependent divider  20 , and a 180° phase shifter  14 . Inputs Beam  1  and Beam  2  are input to 90° hybrid  12 . A first output of hybrid  12  is coupled to the frequency dependent divider  20 . A second output of hybrid  12  is coupled to a middle radiating element  52  of the array  50 . Hybrid  12  may comprise a commercially available wideband 90° hybrids, for example Anaren X3C17-03WS-CT, which as a bandwidth of 690-2700 MHz, with almost constant 90° shift over all frequency band. The 3 dB power dividers,  16 ,  22  may be a multi-section Wilkinson dividers. 
         [0031]    The frequency dependent divider  20  comprises a power divider  22 , a delay line  24 , and a 90° hybrid  26 . The power divider  22  splits the signal from the first output of hybrid  12  into two signals. In this example, the power division of power divider  22  is equal. A first output of the power divider  22  is coupled to a first input of hybrid  26 . A second output of the power divider  22  is coupled to the delay line  24 . The output of delay line  24  is coupled to a second input of hybrid  26 . 
         [0032]    A first output of the frequency dependent divider  20  is coupled to radiating element  51 . In this example, radiating element  51  is the first element in the array  50 . The second output of the frequency dependent divider  20  is coupled to the third element of the array  50 , radiating element  53 , via 180° phase shifter  14 . While 180° phase shifter  14  may be implemented as a discrete component, 180° phase shifter  14  may also be implemented by using a dipole radiator for the radiating element, and alternating the feedpoint relative to the other dipole elements. 
         [0033]    Referring to the frequency dependent divider  20 , the delay line  24  imposes a phase delay to a signal which coupled to the second input of hybrid  26 . However, if the delay line is a fixed length, the phase delay experienced by a signal varies with frequency. That is, for a given fixed time delay, higher-frequency signals experience more phase delay than low frequency signals. Hybrid  26 , therefore, receives equal amplitude signals, where the signals to one input experience increasing phase delay with increasing frequency. Hybrid  26  outputs equal phase, variable amplitude signals, where the amount of amplitude difference increases with increasing frequency. 
         [0034]    With proper selection of delay line  24 , HPBW can be stabilized over a desired frequency band. For example, amplitude ratio between outputs of the frequency dependent divider  20  (and the first and third radiating elements,  51 ,  52 , of array  50 ) may be written as: 
         [0000]        A 1/ A 3=[(1−sinφ)/(1+sinφ)]½
 
         [0035]    Or, P 1 /P 3 =10log [(1−sinφ)/(1+sinφ)] [dB], where φ is electrical length of delay line. When φ=0° or 180°, both elements have the same amplitude. When φ=90° or 270°, one element has 0 amplitude, and another one amplitude 1. 
         [0036]    In the example of  FIG. 1A , the frequency dependent divider  20  is placed between radiating elements  51  and  53 . The delay line is selected to be a regular transmission line with an electrical length of 180° for 1.7 GHz. For other frequencies y=233° for 2.2 GHz; 286° for 2.7 GHz. For both Beam  1  and Beam  2 , beam forming network  10  provides a power distribution at radiating elements  51 ,  52  and  53 , respectively, of 0.7, 1, and 0.7 at 1.7 GHz, 0.36, 1 and 0.88 at 2.2 GHz, and 0.14, 1 and 0.98 at 2.7 GHz. Beam forming network  10  also provides 90° phase differences between radiating elements to create two beams. 
         [0037]    For  FIG. 1B , results of a simulation of radiating patterns for one beam at three frequencies are shown. Spacing between elements for this example is selected at 80 mm and 60 mm. As one can see from  FIG. 1B , HPBW is stabilized to 41+/−3° in this example. 
         [0038]    Another aspect of the present invention is to compensate for the array factor and radiating element patters of a phased-array multi-beam antenna with changing frequency. For example,  FIG. 2A  illustrates a schematic diagram of another example of the present invention. Beam forming network  30  produces two beams via array  60  comprising radiating elements  61 - 64 . Beam Forming Network  30  comprises a 90° hybrid  12   a,  or power divider  16 , a frequency dependent divider  20 , and 180° phase shifter  14 . Hybrid  12   a  may comprise a non-equal 90° hybrid  1  (−3.8 dB, −2.4 dB) for improved sidelobe suppression. 
         [0039]    Beam  1  and Beam  2  signals are input to hybrid  12   a.  A first output of hybrid  12   a  is coupled to power divider  16 . A second output of hybrid  12   a  is coupled to radiating element  63  (the third radiating element of array  60 ). A first output of power divider  16  is coupled to frequency dependent divider  20 . The first and second outputs of frequency dependent divider  20  are coupled to radiating elements  61  and  62  of array  60 , respectively (the first and second elements of array  60 ). A second output of power divider  16  is coupled to 180° phase shifter  14 , which is in turn coupled to radiating element  64 . 
         [0040]    In this example, the power tapering is frequency dependent for radiating elements  61  and  62 , and not frequency dependent for radiating elements  63  and  64 . In particular, in this example, delay line  24  is selected to have φ=270° for 1.7 GHz and 450° for 2.7 GHz. Amplitude and phase distribution is shown in  FIG. 2A  (above the radiating elements). In this example, the amplitude distribution at 1.7 GHz is 0.6, 0, 1, and 0.6 at radiating elements  61 ,  62 ,  63  and  64 , respectively. Similarly, the amplitude distribution at 2.2 GHz is 0.45, 0.42, 1 and 0.6, and the amplitude distribution at 2.7 GHz is 0, 0.6, 1 and 0.6. Accordingly, at 1.7 GHz elements  1 ,  3 ,  4  are radiating, at 2.2 GHz all four elements are radiating, and at 2.7 GHz, elements  2 ,  3 ,  4  are radiating. That is, beam forming network  30 , by reducing amplitude effectively to zero for some radiating elements at certain frequencies, effectively reconfigures Array  60  on a frequency-dependent basis. This feature of frequency re-configurability allows to stabilize beam width and beam position for both beams. 
         [0041]    Calculated radiation patterns are shown in  FIG. 2B  for 1.7, 2.2, 2.7 GHz (for one beam). As one can see from  FIG. 2B , not only is HPBW is stabilized (36+/−4°), but also beam position is stabilized (21.5+/−1.5). 
         [0042]    In another embodiment, all radiating elements of the Array  60  may be continue to be driven, but at different amplitudes at different frequencies. For example, at low frequency of operational bandwidth, central and periphery elements have almost the same amplitude (for example, 0.75; 1; 1; 0.75). For highest frequency of operational bandwidth, periphery elements have much lower amplitude (for example, 0.2; 1; 1; 0.2). 
         [0043]      FIG. 3A  illustrates a schematic diagram of another example of the present invention. Beam forming network  40  produces two beams via array  70  comprising radiating elements  71 - 75 . Beam forming network  40  comprises a 90° hybrid  12   a,  a power divider  16 , two frequency dependent dividers  20 , and two 180° phase shifters  14 . Beam  1  and Beam  2  signals are input to hybrid  12   a.  A first output of hybrid  12   a  is coupled to power divider  16 . A second output of hybrid  12   a  is coupled to radiating element  73  (the third radiating element of array  70 ). A first output of power divider  16  is coupled to a first frequency dependent divider  20 . A second output of power divider  16  is coupled to a second frequency dependent divider  20 . The first and second outputs of the first frequency dependent divider  20  are coupled to radiating elements  71  and  72  of array  70 , respectively (the first and second elements of array  60 ). The first and second outputs of the second frequency dependent divider  20  are coupled to radiating elements  74  and  75  of array  70 , respectively (the fourth and fifth elements of array  70 ). 
         [0044]    The delay line  24  for both frequency dependent dividers  20  in this example comprises a transmission line with φ=283° at 1.7 GHz, 366° at 2.2 GHz, 450° at 2.7 GHz. For example, delay line  24  may comprise 50 Ohm microstrip line with ε=3 and d=85 mm. Spacing between elements for this example is selected 8.0 mm and 60 mm. Hybrid  12   a  may comprise a non-equal 90° hybrid  12   a  (−3.8 dB, −2.4 dB) for improved sidelobe suppression (&lt;−16 dB in this example). 
         [0045]    Amplitude and phase distribution is shown in  FIG. 3A  (above the radiating elements), and calculated patterns are shown in  FIG. 3B  for 1.7, 2.2 and 2.7 GHz (for one beam) and in  FIG. 3C  (for both beams). In this example, at 1.7 GHz, the amplitude distribution is 0.55, 0.25, 1, 0.25, and 0.55 for radiating elements  71 ,  72 ,  73 ,  74 ,  75 , respectively. Similarly, the amplitude distribution at 2.2 GHz is 0.36, 0.45, 1, 0.45 and 0.36, and the amplitude distribution at 2.7 GHz is 0, 0.6, 1, 0.6 and 0. 
         [0046]    For low frequency (1.7 GHz), radiating elements  71 ,  73 , and  75  are handling the most radio frequency energy. For mid-frequency (2.2 GHz), radiating elements  72 ,  73 ,  74  are handling the most radio frequency energy. For high frequency (2.7 GHz), radiating elements  72 ,  73  and  75  are handling all of the radio frequency energy, and radiating elements  71  and  75  are not radiating. Accordingly, this is another example of a frequency reconfigurable antenna. By selectively using different radiating elements at different frequencies, the effective aperture of array  70  is changing proportional to wavelength for both beams, providing almost constant HPBW and beam position angle. 
         [0047]    The example of  FIG. 3A  is more complicated relative to the examples of  FIG. 1A  and  FIG. 2A , but constant HPBW over wide (50%) bandwidth, constant beam position and low sidelobe level constitutes an improvement over the prior art. For example, HPBW and beam position are very stable: 34+/−2° and 21+/−1°, respectively. The beam width and beam position tolerances are about 10 times better than in prior art (U.S. Pat. No. 8,237,619, FIG. 2, FIG. 3: HPBW=40+/−7° and beam position 20+/−4° for twice less frequency band, 1.7-2.2 GHz only with high sidelobes (−9 dB). 
         [0048]    In the above examples, beam forming networks were shown for 1.7-2.7 GHz, but the invention is not limited to this frequency band and may be implemented using any other frequencies. 
         [0049]    Various delay line  24  may be advantageous in various applications. For example,  FIGS. 4A ,  4 B and  4 C illustrate three different variations of delay lines.  FIG. 4A  illustrates a regular transmission line, where phase delay is directly proportional to frequency. For example, it can be  50  Ohm microstrip line with length d, and φ=[2π(εe½)]/λ, where εe—effective dielectric constant of substrate material. This line may be used for the example illustrated in  FIG. 1A  (φ=180°  1 . 7 ) GHz and for  FIG. 3  (φ=283° GHz 1.7). 
         [0050]      FIG. 4B  illustrates a delay line comprising a regular transmission line combined with a Shiffman phase shifter. Because a Shiffman phase shifter provides constant phase over frequency band, phase for this delay line will change more slowly with frequency compared to a regular transmission line as illustrated in  FIG. 4A . 
         [0051]      FIG. 4C  shows a loaded line, where narrow sections (series inductances) are combined with wide sections (parallel capacitances), providing 15-30% “faster” phase change compare to regular line. The example of  FIG. 4C  may be used, for example, to create amplitude distribution shown in the example of  FIG. 2A : φ(1.7 GHz)=270° and φ(2.7 GHz)=450°. 
         [0052]    By using different style of delay line, it is possible to provide desirable dependence of amplitude distribution against frequency, and get optimization of beam position, beam width and sidelobes. 
         [0053]    Wideband 180° phase shifter  14  may be easy realized with dipole radiator, by alternating feedpoint (for 0°, central conductor of feed line is connected to 1 st  dipole arm, for 180° , it is connected to 2 nd  dipole arm). This solution provides constant 180° phase shift over whole frequency band without having to add a discrete phase shifter component. 
         [0054]    In a full array, a combination of 3, 4 and 5 element linear arrays as described in  FIGS. 1A ,  2 A,  3 A may be implemented. Such combinations may be advantageously used to reduce azimuth sidelobe level. For example,  FIGS. 5A and 5B  illustrate alternating five element arrays  70  and four element arrays  60 .  FIGS. 5C and 5D  illustrate alternating pairs of five element arrays  70  and four element arrays  60 . Also, for further sidelobes&#39; and beam shape optimization, horizontal spacing between elements can also be different. Examples shown in  FIG. 5  are related to antennas with +/−45 degree dual polarization (most common in base station antenna technology), but of course the same solutions can be applied to antennas with any single or dual polarization.