Abstract:
In a preferred embodiment, a system of physical measurement and display of a parameter, including: accepting an input signal representative of a value of the parameter, the input signal having an arbitrary shape and amplitude; and displaying the value of the parameter in an arbitrary shape; including the use of analog and digital circuitry and nonvolatile memory.

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     U.S. application Ser. No. 08/633,423 is a continuation-in-part of application Ser. No. 08/423,784, filed Apr. 18, 1995, and titled COMPACT, LOW-COST, SEMICONDUCTOR DEVICE FOR RECEIVING ARBITRARY INPUT PARAMETERS AND DRIVING SELECTED DISPLAY DEVICES, AND METHODS, now abandoned. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to the display of measured parameters generally and, more particularly, but not by way of limitation, to novel compact, low-cost, semiconductor device and methods for receiving arbitrary input parameters and driving selected display devices and methods therefor, which are particularly useful in motorized vehicles. 
     2. Background Art 
     Particularly in motorized vehicles, but not necessarily limited thereto, there is need for measuring and displaying physical parameters such as pressure, temperature, liquid level, voltage, and current and, in some cases, providing high and/or low indications thereof. Conventionally, almost all of these functions are satisfied by various electromechanical gauges which are relatively expensive and space-consuming. When high and/or low indication is required, auxiliary sensors and displays are frequently employed, as it is more economical to do so rather than derive the presence of an extreme condition from the normal measurement signal. Also, in order to accommodate the electromechanical display devices, the sensors must often be designed to be very non-linear, since the amount of freedom to shape the display device is quite limited. This latter factor introduces additional cost and space consumption to conventional systems. 
     It would be desirable to have a method and apparatus for physical parameter measurement and display which is particularly well suited for parameters such as pressure, temperature, liquid level, voltage, and current. There exists a wealth of inexpensive sensors and transducers which can provide electrical inputs to such a device. Unfortunately these transducers and sensors come in a very large number of electrical forms and non-linearities. It is, therefore, a first objective of the present invention to accept an arbitrary shape and, to some extent, arbitrary amplitude input signal. 
     It is very often desirable to contour the displayed information in order to shape, compress, or expand the parameter for ergonomic or resolution reasons. It is, therefore, a second objective of the invention to display the parameter in an arbitrary shape. 
     Many of the available inexpensive transducers and sensors have rather low impedances, requiring an excessive amount of power (in the context of the “micro-instrument” provided by the present invention). It is, therefore, a third objective of the invention to operate the mating transducer in one of several power conserving modes. 
     Some transducers are designed to provide an increasing output for an increasing physical parameter while others produce a decreasing output for an increasing input. It is, therefore, a fourth objective of the invention to treat the input in a normal (ascending) mode or an inverted (descending) mode. 
     In some cases the input signal comes from a calibrated source (e.g., absolute volts), but in other cases the input signal comes from a transducer which must be excited by the device (frequently in a power conserving mode). It is, therefore, a fifth objective of the invention to operate in either an “absolute” or “ratio” mode. 
     Some parameters are inherently jittery, or noisy (such as a fuel sender in a sloshing tank). It is, therefore, a sixth objective of the invention to provide varying degrees of damping or slew rate control. For example, relatively fast for pressure and voltage, or relatively slow for fuel level and temperature. 
     In critical applications, a second sensor is frequently introduced, as noted above, to warn or shut down an operation when an out-of-specification condition is detected. It is a seventh objective of the invention to provide a programmable over and/or under warning output. 
     In other critical applications, both a second and third sensor may be introduced to warn and/or shut down an operation when either a low or high out-of-specification condition is detected. It is, therefore, an eighth objective of this device to provide at least one additional programmable over and/or under warning output. 
     Most instruments designed for the vehicle market incorporate illumination. It is, therefore, a ninth objective of the invention to provide self-illumination by directly driving light-emitting diodes. 
     The light emitting diodes may be illuminated sequentially (pointer mode) or additively (bar graph mode). It is, therefore, a tenth objective of the invention to provide self-illumination in either of two modes, pointer or bar graph. 
     In some applications, such as portable battery operated equipment, it becomes necessary to run the device and transducer at very low power levels. It is anticipated that an inherently low power (or high impedance) transducer will be operated in one of the power conserving modes, and that the device will be operated in a direct drive LCD mode. In this mode a liquid crystal display may be driven without the need for temperature compensation over the temperature range of −40 degree C. to +85 degree C. It is, therefore, an eleventh objective of the invention to directly drive a liquid crystal display. 
     The LCD may be activated sequentially (pointer mode) or additively (bar graph mode). It is, therefore, a twelfth objective of the invention to provide low power LCD activation in either of two modes, pointer or bar graph. 
     Both the LED and LCD Display drives address individual segments. These segments may be arranged in limitless configurations; straight line vertical, straight line horizontal, straight line diagonal, curved up, curved down, patterns, matrices, etc. Two or more light emitting diodes can be simultaneously energized in series. Two or more LCD segments can be simultaneously energized in parallel. Static light emitting diodes can be energized directly, but static LCD segments require an AC drive signal. The static elements are envisaged as part of the scale legend or information, and are intended to be active whenever the power is applied. It is, therefore, a thirteenth objective of the invention to provide a continuously operating LCD segment driver. 
     The light emitting diodes can be assembled in specific color coded formats. For example, the traditional OK-Caution-Warning/Stop sequence of Green-Yellow-Red can be arranged within the display to impart that interpretation, while simultaneously “pointing” to the measured value. When coupled with scale expansion and contraction contouring the display can impart an unusually large amount of accurate information with relatively few display elements. It is, therefore, a fourteenth objective of the invention to provide color coded status information integral with the measurement. 
     When an out-of-specification condition is detected, it is often desirable to “flash” the display to draw attention to the fact that a problem exists. It is, therefore, a fifteenth objective of the invention to provide various flashing modes at the upper limits of the range or the lower limits of the range. 
     If an out-of-specification condition can occur at both extremes of the range, it may be desirable to “flash” the display for both conditions. It is, therefore, a sixteenth objective of the invention to provide various independent flashing modes at both the upper limits of the range and the lower limits of the range. 
     When information is known to vary at a slow rate, it may be desirable to limit the update of the display to rather long intervals to reduce apparent “flicker” when a borderline reading is made in a noisy environment. It is, therefore, a seventeenth objective of the invention to provide various conversion rate intervals. This will usually be used in conjunction with the slew rate control (objective six). 
     When information is known to be somewhat noisy, it may be desirable to introduce hysteresis to lessen the likelihood of borderline flicker. It is, therefore, an eighteenth objective of the invention to provide an hysteresis on/off feature. This will usually be used in conjunction with the conversion rate control (objective seventeen) and the slew rate control (objective six). 
     Having separate control over slew rate, conversion rate, and hysteresis allows for a large number of possibilities in solving the jitter and noise problems in systems with various response time requirements. Taken together they allow for ergonomic fine-tuning relative to the display, and/or warning response fine-tuning relative to the output signals. 
     The method and apparatus described herein has been architecturally optimized for a singe chip triple technology implementation (triple tech=analog, digital, and non-volatile memory in complementary MOS), on a particularly small and inexpensive die. The solution for the complete instrument must be cost competitive with the optimized evolved electromechanical gauges and their one hundred year market domination. It is, therefore, a nineteenth objective of the invention to be substantially integratable in a small, inexpensive, triple technology chip. 
     It is intended that one and only one chip serve all the instrumentation needs of a product line that will operate from numerous power sources, with innumerable transducers, displaying numerous transforms in either LED or LCD formats, with many functional options. Therefore, it is desirable that the invention will be programmable at the chip level via non-volatile memory, and at the circuit board level via “stuffing” instructions. Typical stuffing instructions will be resistor values for voltage options. Resistor values for classes of transducers, capacitor values for filter response, LED arrays for specific color formats, LCD assemblies for specific legend, etc. It is, therefore, a twentieth objective of the invention that all of the integratable options be programmed in non-volatile memory at the time of instrument manufacture. It should be noted that the vast percentage of applications will be satisfied with relatively few combinations of voltage, LED color assignment, and transducer “class”. Therefore, very high level subassemblies could be stocked waiting for programming information that transforms them into any of thousands of option combinations. 
     The possibilities of transducer calibration for, say, resistive senders is huge, when accounting for absolute value, offsets, linearity, and direction (increasing versus decreasing). However, these can be conveniently lumped into broad classes of absolute value, say, two or three broad classes. When this is done via resistor stuffing at the subassembly level, it becomes a simple matter to fine tune the absolute value, adjust the offset, resolve the linearity relative to the desired display response, and assign the direction in non-volatile programming at this essentially “functionally complete” level of assembly. 
     Further elaborating on this “one chip for all” theme, it is the intention in this design to optimize the low power consumption of the chip without compromising the high current drive capability, or the size (cost) of the chip. It is, therefore, the twenty first objective of the invention to operate at a low quiescent current, while maintaining high current drive capability. 
     There are two areas of absolute reference on-chip generation that require discussion. The first is the time base, and the second is the voltage reference. A very low cost, but moderately accurate time base is needed to establish all internal clocking, analog-to-digital conversion, LCD drive waveforms, display flashers, non-volatile memory loading, non-volatile memory viewing, etc. A fully integrated R-C oscillator operating at 131 kiloHerz (after trimming) followed by binary counting chains conveniently produces 64 Herz for the LCD drivers and 2 Herz for the flashers (divided by 2 11  and divided by 2 16 , respectively). A low cost band gap voltage regulator can be created in CMOS with fair temperature coefficients and curvature error, but poor absolute value. With trimming, the regulator error can be brought into acceptable limits. 
     It is, therefore, a twenty second objective of the invention to integrate a small, low-cost oscillator and trim this oscillator with instructions from the non-volatile memory, with these instructions initially inserted during wafer test, but re-written during final assembly, if required. 
     It is, further, a twenty third objective of the invention to integrate an efficient, small, low-cost band gap regulator and trim the regulator with instructions from the non-volatile memory, with these instructions initially inserted during wafer test, but re-written during final assembly, if required. 
     Other objects of the present invention, as well as particular features, elements, and advantages thereof, will be elucidated in, or be apparent from, the following description and the accompanying drawing figures. 
     This disclosure describes a display with ten (10) distinct states producing pointers, bar graphs, pictograms, icons, etc. The method can apply to any number of states, with 12, 15, 20, 25, etc., being quite practical. In every instance, the memorized thresholds will be one less than the number of display states. For example, 9 thresholds for 10 states and 24 thresholds for 25 states. It is, therefore, the twenty fourth objective of the invention to store in non-volatile, memory during the time of instrument manufacture a number of display states. Each of these thresholds will be stored digitally with a resolution commensurate with the desired accuracy and resolution of the analog-to-digital conversion process. It is, therefore, the twenty fifth objective of the invention to perform an analog-to-digital conversion process of sufficient accuracy and resolution to satisfy the stated instrument accuracy. Most often, the conversion process is expected to have resolutions of 6, 7, 8, 9, or 10 bits. This disclosure will focus on 8 bit examples, as they are quite appropriate for instrument applications in automotive and off-highway vehicles, motor/generator sets, motor/compressor sets, etc. 
     SUMMARY OF THE INVENTION 
     The present invention achieves the above objects, among others, by providing, in a preferred embodiment, a system of physical measurement and display of a parameter, comprising: accepting an input signal representative of a value of said parameter, said input signal having an arbitrary shape and amplitude; and displaying said value of said parameter in an arbitrary shape. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Understanding of the present invention and the various aspects thereof will be facilitated by reference to the accompanying drawing figures, submitted for purposes of illustration only and not intended to define the scope of the invention, on which: 
     FIGS. 1-3 are block/schematic diagrams showing alternative analog-to-digital conversion circuits useful in the present invention. 
     FIGS. 4A-4C are fragmentary schematic diagrams showing a method of changing the range of an input voltage signal. 
     FIGS. 5-7 are block/schematic diagrams showing additional alternative analog-to-digital conversion circuits useful in the present invention. 
     FIGS. 8A-8C are block/schematic diagrams showing the circuitry of the semiconductor device of the present invention. 
     FIG. 8D shows the arrangement of FIGS. 8A-8C for the reading thereof. 
     FIGS. 9-11 are block/schematic diagrams showing the device of FIG. 8 used with alternative display elements and with alternative input and out-of-tolerance indication arrangements. 
     FIG. 12 is a fragmentary block/schematic diagram showing an alternative out-of-tolerance arrangement. 
     FIG. 13 is table setting forth a programming and calibration memory map for the device of FIG.  8 . 
     FIG. 14 illustrates a typical timing sequence for the analog-to-digital conversion circuit of the semiconductor device. 
     FIGS. 15-17 are block/schematic diagrams showing alternative auxiliary elements used with the semiconductor device. 
     FIG. 18 is a block/schematic diagram showing input switching circuitry for use when the present invention is in a power conservation mode. 
     FIG. 19 is an output driver truth table for the semiconductor device. 
     FIGS. 20-22 are schematic diagrams showing output driver arrangements for use with various display elements. 
     FIG. 23 comprises waveforms showing the “ON” and “OFF” states of a segment (pointer) driver and the “ON” state of a legend driver relative to a back plane driver. 
     FIG. 24 is an upper flasher truth table for the semiconductor device. 
     FIG. 25 is a lower flasher truth table for the semiconductor device. 
     FIG. 26 is a “HI OUT” truth table for the semiconductor device. 
     FIG. 27 is a “LO OUT” truth table for the semiconductor device. 
     FIG. 28 shows the resistance/pressure relationship of a non-linear resistive transducer. 
     FIG. 29 shows the resistance/temperature relationship of a two non-linear resistive transducer. 
     FIG. 30 is a schematic diagram of the input circuitry associated with the transducer of FIG.  28 . 
     FIG. 31 is a schematic diagram of the input circuitry associated with the transducer of FIG.  29 . 
     FIG. 32 is a table presenting a summary of the complete transducer-to-output transform for the input described by FIGS. 28 and 30. 
     FIG. 33 is a table presenting a summary of the complete transducer-to-output transform for the input described by FIGS. 29 and 31. 
     FIG. 34 is a table presenting a memory map for the example of FIGS. 28,  30  and  32 . 
     FIG. 35 is a top plan view of a single-chip CMOS implementation of the present invention. 
     FIGS. 36-38 are isometric, top plan, and side elevational views, respectively, of an instrument package incorporating the embodiment of the present invention shown on FIG.  9 . 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Reference should now be made to the drawing figures, on which similar or identical elements are given consistent identifying numerals throughout the various figures thereof, and on which parenthetical references to figure numbers direct the reader to the view(s) on which the element(s) being described is (are) best seen, although the element(s) may be seen also on other views. 
     FIG. 1 illustrates an analog-to-digital converter, generally indicated by the reference numeral  100 , which is employed as a participant in the filtering process in the present invention. Here, a voltage-to-frequency converter  102  operating over an input range of 0 to 2 volts and an output range of 0 to 2 kiloHerz followed by a 12-stage binary counter  104  is an excellent integrating analog-to-digital converter over, say a 2 second measurement interval, with 0 to 2 volts=0 to 2 kiloHerz=0 to 4000 counts over 2 seconds. This results in a 0 to 250 total in an output register  106  coupled to binary counter  104 . Every 2 seconds, a transfer command is issued by transfer and reset circuitry  108  followed by a reset command. These commands are infinitesimal in duration relative to the integration interval (microseconds versus seconds). The interval is easily expanded or contracted, as well as the voltage-to-frequency conversion factor and the countdown stages, to produce very practical conversion (integration) intervals from tenths of seconds to hundreds of seconds. For example; an input range of 0 to 2.5 volts produces a voltage-to-frequency output of 0 to 1.25 kiloHerz followed by a 17-stage binary counter yielding the following states. 0 to 2.5 volts=0 to 1.25 kiloHerz=0 to 125,000 counts over 100 seconds. This results in a 0 to 244 total in the output register. 
     The integration over a predetermined interval may also be performed, as illustrated on FIG. 2, in the analog front end inherent within a dual slope analog-to-digital converter, generally indicated by the reference numeral  120 . In this case the input voltage is integrated with respect to time over the desired interval within an analog integrator  122 , and is then quickly de-integrated under time and counting control. Since the typical conversion will be on the order of 8 bits and the integrating interval will usually be on the order of seconds, the conversion can be kept simple by merely allowing the de-integrate current to overwhelm the input signal over an interval of, say, 1 millisecond, without concern for switching the input, or for creating any significant error. During the de-integrate interval, integrator  122  is continuously “up”-integrating in direct response to the input signal. When the time to convert occurs, say every 2 seconds, a reset command (R) is issued by reset and convert circuitry  124  to an output counter and register  126 , followed immediately by a convert command (C) which sets a latch  128 . Q of latch  128  goes to a logical “1” which activates the de-integrate current source  130  and enables the 250 kiloHerz clock (CLK) which will advance counter  126  through AND gate  132  over the 1 millisecond de-integrate interval required by a maximum input signal. A half-scale input would require 500 microseconds, yielding 125 counts, and so forth. When the output of integrator  122  reaches a “base-line” level a baseline detector  134  changes state, immediately resetting latch  128 , thus stopping the flow of the de-integrate current and the counting in output counter/register  126 . Baseline detector  128  may be conveniently set at 0.1 volt for the single positive power source example shown. In a plus-and-minus power supply system base line detector  134  may be conveniently set at 0.0 volts thus becoming a zero crossing detector. 
     The de-integrate current source  130  for integrator  122  is shown floating in analog-to-digital converter  120  on FIG.  2 . That current source may be fixed against a power supply related reference by translating the input signal to a positive baseline in a single power supply system, thus allowing for bidirectional current flow into the integrator. For example; a normal 0-to-2-volt input signal can be translated to 1-to-3 volts and the input integrator and de-integrate current source can be modified simply as shown on FIG. 3 in which an analog-to-digital converter, generally indicated by the reference numeral  140  includes the same elements as analog-to-digital converter  120  shown on FIG. 2, with the addition of a +1.00-volt reference  142 . 
     The following calculations illustrate some practical values for analog-to-digital converter  140  (FIG.  3 ). The assumptions are: the basic circuit power is +6.00 volts; the +1.00-volt reference is derived directly from the +6.00 volts; the input signal ranges from +1 to +3 volts; the conversion interval is 2.00 seconds; the integrating capacitor CINT is 1.00 microfarad; the range on the output of integrator  122  with the maximum input of 3 volts is 1 to 5 volts; the 1.00-volt lower limit is (arbitrarily) governed by baseline detector  134  with its 1.00-volt reference input; the 5-volt upper limit is controlled by component values; and de-integrate time=1 millisecond maximum. 
     (1) Q=CΔE=1×10 6 (5−1)=4 micro-Ampere-seconds. 
     Integrator input current @ max. input voltage equals (−4 micro-Ampere-seconds)/2 seconds=−2 micro-Ampere. 
     (3) R INT =(1-3 volts)/(−2 micro-Amperes)=−2/−2 megOhms=1 megOhm. 
     (4) Integrator input current required to discharge the capacitor by 4 volts in 1 millisecond equals (+4 micro-Ampere-seconds)/0.001 second)=+4000 micro-Amperes=+4 milli-Amperes (See (6)). 
     (5) R DIN =(6−1)/4 kilo-Ohms=1.25 kilo-Ohms (See (6)). 
     (6) Theoretically the de-integrate current must provide the 4 micro-Ampere-seconds discharge plus the (assumed) negligible input charge of 2 micro-Amperes for 1 millisecond=2 nano-Ampere-seconds. (4) and (5) then become 4.002 milli-Amperes and 1.24938 kilo-Ohms, respectively. 
     The following fragmentary schematics show one form of a simple translation from a 0-to-2-volt input (FIG. 4A) into a 1-to-3-volt input (FIG. 4B) for a 0-to-1000-Ohm resistive transducer, using the calculations indicated on FIG.  4 C. 
     If plus and minus power supplies are used, the integrator can be further simplified, especially if used with negative input signals. In its simplest form, it becomes the circuit shown below, generally indicated by the reference numeral  150 , on FIG. 5, where R INT  has been moved to the input and a zero crossing detector  152  replaces baseline detector  134  (FIGS.  2  and  3 ). 
     Usually, however, the input signals will be positive (and small, in this case) relative to common, and an inverting amplifier will be used ahead of the integrator. This is shown on FIG. 6 where an analog-to-digital converter, generally indicated by the reference numeral  160  includes an inverting voltage amplifier  162  in front of integrator  122 . This can be advantageous when dealing with low level signals, where the input offset voltage performance of input amplifier  162  is best separated from the input offset current performance of integrator  122  (via the input amplifier&#39;s voltage gain). 
     When the analog-to-digital converter is not a major participant in the filtering process, it can be simplified to become a digital-to-analog converter and analog comparator as shown in the circuitry illustrated on FIG.  7  and generally indicated by the reference numeral  170 . The advantages are: 
     (1) Easy implementation in single chip integrated circuits. 
     (2) Relatively small circuit area. 
     (3) Independent of frequency error (depends on voltage REF only). 
     The disadvantage is, of course, that no integration (or filtering) is inherent in the conversion process and circuitry  170  is always intended to be used in conjunction with a filtered input. 
     When a conversion is requested (typically ½, 1, 2, 4, etc., seconds) a short reset command (R) from a reset and convert circuitry  172  resets an output counter,  174  and is followed immediately by a convert command (C) which sets latch  176 . The set latch  176  enables a relatively fast clock (e.g. 131 kiloHerz) via AND gate  178 . The counter counts up until the output from a digital-to-analog converter  180  exceeds the output from a low pass filter  182 , at which time a comparator  184  changes state, resetting latch  176 , and disabling the clock. The information in output counter  174  remains static until the next conversion interval, and it therefore serves as the output register for circuitry  170 . The maximum conversion time for the example shown is 255×7.6 microseconds=1.95 milliseconds. Meantime, the response of filter  182  could be several seconds (e.g., 1 megOhm and 2 microfarads=2 seconds), while the conversion rate could be 4 seconds, etc. Therefore,the conversion time could occupy less than 0.05% of the “update” period, eliminating the need for a separate output register. Counter  174  limits at 255 to prevent overflow. 
     Further simplification is possible by combining at least a portion of the level decoder with the digital-to-analog conversion process. If the digital-to-analog converter converts the sequential representation of the thresholds stored in memory, rather than an independent conversion, it could eliminate the need for the output counter and register and related clock and reset. When a conversion is required, the memory is read row-by-row, and is converted row-by-row, until the conversion is greater than (or less than, depending upon normal or invert selection) the filtered input signal. This level is then held until the next conversion is required. 
     FIG. 8 illustrates the circuitry on the semiconductor device of the present invention, generally indicated by the reference numeral  200 . Device  200  includes the analog-to-digital conversion circuitry  170  (FIG. 7) which received an input signal from a transducer (not shown) at “XDCR IN” through an analog buffer  202  and an integrated switched resistor array  204 . The output of circuitry  170  is fed to a level decoder  210  coupled to a drive mode generator  212  which provides display drive signals at a plurality of outputs, as at  214 . Also on device  200  is a 96-bit non-volatile memory  220  the functions of which will be described below. 
     Illustrating the universality of semiconductor device  200 , FIG. 9 shows the device combined with auxiliary circuitry to produce an instrument, generally indicated by the reference numeral  230 , with a resistive transducer input, the transducer being represented by R x . Device  200  drives a 10-segment LED display  232  in the pointer mode, and each of the warning outputs drives remote LEDs  234  and  236  in the pull-down mode. Power dissipation resistors R 4 A and R 4 B may be provided as a single, although slightly larger, resistor. FIG. 10 shows semiconductor device  200  combined with auxiliary circuitry to produce an instrument, generally indicated by the reference numeral  240 , with a voltage input. Device  200  drives a 10 segment LED display in the bar graph mode, and each of warning outputs drive remove LEDs  244  and  246  in the pull-up mode. FIG. 11 shows semiconductor device  200  combined with auxiliary circuitry to produce an instrument, generally indicated by the reference numeral  250  with an offset zero resistive input, the offset being provided by resistor R y . Device  200  drives a 10-segment plus legend LCD display  252  in either the printer or bar graph mode. The warning outputs of device  200  drive remote low power relays  254  and  256  in the pull-down mode, signaling separate high and low conditions. FIG. 12 shows an LED  260  “OR&#39;d” in the pull-down mode, via diodes  262  and  264 , with the single LED signifying an out-of-tolerance condition (high or low). This can be used in conjunction with 10-segment LED or LCD displays. The particular combinations of inputs, displays, and warning signals of instruments  230 ,  240 , and  250  are for illustrative purposes only and any combination of the elements shown, as well as others may be provided within the intent of the present invention. 
     Referring back now to FIG.  8  and FIG. 13 together, the latter setting forth a programming and calibration memory map, an integrated oscillator  270  is calibrated during test by inserting one of eight trim possibilities into memory locations CAL 4 , CAL 5 , and CAL 6 . These control resistor and/or capacitor arrays in oscillator  270 . The nominal 131-kiloHerz drives a long binary counter chain  272  which generates all of the system&#39;s sub clocks. The chain may proceed to even lower frequencies than the ¼ Herz (4 seconds) illustrated. 
     An integrated band gap voltage reference  280  is also calibrated during test by inserting one of eight trim possibilities via memory locations CAL 1 , CAL 2 , and CAL 3 . These control resistor arrays in voltage reference  280 . The output from the reference is buffered by an analog amplifier  282  for internal and (on occasion) external use. Switch  284  at the output is under control of memory location PR 23 . When PR 23  is a logical “1” the switch is closed and the semiconductor device is in the “absolute” mode where the reference to converter  170  is the trimmed absolute band gap reference. The signal present at “REF IO” is now an output and may be used as an auxiliary reference elsewhere within the completed instrument or system. When PR 23  is a logical “0”, switch  284  is open and device  200  is in the “ratio” mode where the reference to the converter  170  is provided from an external source. 
     The “XDCR IN” signal is buffered and filtered by analog buffer amplifier  202  and its output R-C low pass filter. Resistor array  204  portion is an integrated switched resistor array controlled by memory locations PR 24  and PR 25 , allowing for four resistor possibilities. The capacitor portion is completed externally, and usually represents major jumps in time constants, while the resistors produce smaller steps (typically 15- or 20-to-1 for capacitor and 2-to-1 for resistor). 
     Non-volatile memory array  220  is a 96-bit (12 rows×8 columns) structure, rows  1  thru  9  store the nine thresholds, T 11  thru T 99 , as 8-bit words to accommodate individual decimal numbers from 0 to 255. Rows  10  and  11  store individual programming bits or clusters of bits. Each bit is identified as PR 1  followed by  1  thru  8 , or as PR 2  followed by  1  thru  8 . Row  12  stores two clusters of calibration data (CAL 1  thru CAL 3 , CAL 4  thru CAL 6 ) and two programming bits, CAL 7  and CAL 8 . FIG. 13 summarizes the programming and calibration allocations. Each of these memory locations appear twice within the block diagram. First, within the memory array itself, and second at the function control location. 
     Data is loaded into memory  220  and examined coming out of memory via the bidirectional serial data port, PIO,  300 . Each group of 8 data bits is accompanied by a 4-bit address in order to identify the related memory row. Three inputs and one output are involved in writing new data into the memory. PCL (programming clock) is the output and is derived from the main counter chain (C 0  thru C 19 ). The choice of frequency is somewhat arbitrary, but it is necessary as a synchronizer for the three inputs. Typically CX might be C 6 , C 7 , or C 8 . PEN is the programming enable input. It puts the PIO port into the input mode and enables a programming sequencer  302  when presented with a logical “1” PRV is the programming voltage input. The programming (and re-programming) of memory  220  requires voltages outside of the normal power supply regime. In order to maintain maximum cost effectiveness for semiconductor device  200 , the normal on-chip programming voltage generation is moved off-board; especially beneficial in light of the fact that programming may often be a one-time manufacturing operation. 
     Programming sequencer  302  moves through an orderly erase and write sequence under timekeeping from clock CY which, of course, is related to CX. It issues instructions to erase, write, and advance through the rows, via time and voltage control. 
     Conversion rate is programmed for one of four rates depending upon the application&#39;s needs. Memory locations PR 26  and PR 27  control the selection. In this example the rates may be ½ second, 1 second, 2 seconds, or 4 seconds. 
     For longer practical conversion times which may be employed, i.e., 8-16 seconds, the thermal time constants for all components are such that effects of temperature are insignificant. The time constants are on the order of minutes, so the time averaging scheme makes any temperature ripple due to thermal effects virtually unnoticeable. 
     Control over conversion rate is usually dictated by ergonomic and/or signal noise/jitter considerations; however, it can be functionally extended to assist in transducer power conservation. In the extreme case of a 0-to-100-Ohm transducer in series with a 100-Ohm bridge resistor with 5-volt bridge excitation, the worst case current will be 50 milli-Amperes. If this current must be supplied within the instrument from, say, 15 volts (high float charge voltage of a 12-volt DC system) the maximum power will be 750 milliwatts. This may be beyond the thermal capacity of a “micro-instrument”. If this power could be modulated to ½ or even ⅛, the time-averaged transducer excitation load would reduce to 375 or 94 milliwatts, respectively. PR 28  controls the conversion width modulation (CWM) output from a conversion rate selector  310  to energize a transducer just before measurement, thereby to conserve power. When PR 28  is a logical “1”, the CWM output is active for the last ½ of the conversion period. When PR 28  is a logical “0” the CWM output is active for the last ⅛ of the conversion period. FIG. 14 shows a typical timing sequence (on a variable time base scale) for a 4-second conversion with ⅛ duty cycle (PR 28 =0). 
     Continuing to refer to FIGS. 8A-C, the CWM signal is then buffered and presented externally as XPM (transducer power modulation). Typically, this will be used in applications where power conservation is mandatory such as: high ambient temperature; very low transducer resistance; heat dissipation restrictions in “micro-instruments”; portable battery operation; etc. Generally, this will be used with relatively fast filter time constants, because the signal must achieve approximate stabilization within the pulse&#39;s width. In this example, a maximum time constant on the order of 90 milliseconds is recommended to keep the error under around ½% or 70 milliseconds to keep the error under around {fraction (1/10)}%. The circuit is completed by using an external buffer amplifier  320  as shown in FIG.  15 . To further improve power conservation, the output from XPM is divided down by the resistor pair  330 / 332 . This also allows “headroom” for buffer amplifier  202  to operate when powered from the same supply voltage (+DC) as semiconductor device  200 . FIG. 16 shows another variant where XPM is inverted and directly drives an external low resistance P channel FET  340 , thus allowing drive of a grounded resistive transducer  342  with only one external component in a ⅛ duty cycle power conserving mode. On both FIGS. 15 and 16, the loads have been removed from “HI OUT” and “LO OUT” and a liquid crystal display is used for maximum reduction in power consumption. 
     On both FIGS. 15 and 16 a series regulator  350  replaces the simple Zener diode shunt regulator in order to realize the gains achievable with the transducer power modulation (by allowing the total input current to drop during the modulator “off” time). 
     FIG. 17 shows a low power shunt regulator alternative with a relatively high current emitter follower transducer driver and N channel MOSFET lamp drivers  360  and  362 . This configuration allows a rather low power “micro-instrument” to drive low impedance off-board loads, such as incandescent lamps  364  and  366 . All high current paths draw directly from unregulated power. 
     FIG. 18 illustrates an implementation in which, when operating in the power conservation mode, it is possible to maintain relatively long filter response times while maintaining relatively decent accuracy by operating the filter in a track-and-hold manner. If, for example, we assume that the maximum sample-to-sample parameter change is on the order of 5%, the settling time to get within ½% will be 2.3 time constants rather than the 5.3 time constants required for a 100% step change (e −2.3  around 0.10,×5%=0.5%; e −5.3  around 0.005,×100%=0.5%). Therefore, the filter could be 2.3 times longer (slower by 5.3/2.3=2.3) for the same effective accuracy. This modification can be implemented by inserting a fifth state into the integrated switched resistor array  204  which is under control of the CWM signal. When CWM is low, the resistor switches are forced open, thus eliminating the discharge path back through the output of analog buffer amplifier  202 . The input impedance of the analog comparator is assumed to be infinite. THI is the track/hold input. It is jumpered to +DC for full time tracking or to XPM for hold during CWM low and track during CWM high. This could be programmed in non-volatile memory  220 . The other elements of digital-to-analog converter  170  operate as previously described. 
     Continuing to refer to FIGS. 8A-C, level decoder  210  compares the converted data to the nine stored thresholds (T 11  thru T 99  in memory  220 ) and places that data within one of ten possibilities. These possibilities are a function of: firstly, the nine stored thresholds; secondly, whether or not there is hysteresis, and thirdly, whether the input function is normal or inverted. CAL  8  controls hysteresis, with a logical “1” commanding hysteresis to be active. CAL  7  controls the normal/invert functions, with logical “1” indicating the normal mode. When hysteresis is “on”, the threshold is shifted down by one bit after passing up through the threshold in order to prevent dithering when the input signal resides at that threshold. Alternatively, this function could be incorporated within conversion circuitry  170 . When hysteresis is “off,” no threshold shift occurs. 
     In the normal mode, the L 00  output is controlled by converted values less than T 11 , whereas in the inverted mode it is controlled by converted values greater than T 11 . at the other end of the scale a similar inversion occurs, where in the normal mode the L 99  output is controlled by converted values equal to or greater than T 99 , but, in the inverted mode, it is controlled by converted values equal to or less than T 99 . FIG. 19 is an output driver truth table which illustrates the normal/invert function relative to the display functions of a pointer/bar graph. The example shows a linear 0-to-(+X) volts function which is the normal mode, and a linear (+X)-to-0 volts function which is the inverted mode. The range is assumed to comprise 10 evenly spaced converted values of 25 each for a total of 250. 
     With respect to zero protection limit and positive protection limit: There are physical protection limits at “XDCR IN” which prevent damage to semiconductor device  200  in the event of input overshoot or undershoot, provided the input current is limited to a sub latchup value (typically 10 to 100 MA depending upon the input protection design). In addition, there are overflow and underflow limits within analog-to-digital converter  170  to prevent the creation of illegal numbers (due to rollovers) outside of the range of 0 to 255. 
     Drive mode generator  212  conditions the 10 zones of decoded data into forms needed for the 10 LCD or LED drivers. Memory location PR 21  selects the LED or LCD mode, and memory location PR 22  selects the pointer or bargraph mode. C 11  provides the 64-Herz reference for generating the direct drive LCD waveforms. FIG. 20 shows the output drivers operating an LED pointer display, FIG. 21 shows the drivers operating an LED bargraph display, and FIG. 22 shows the drivers and their connection destinations to a 10 segment plus legend LCD display. FIG. 23 shows the LCD waveforms for the ever-present legend driver and a typical segment in its “on” and “off” states relative to the ever-present back plane driver. In the pointer mode, one and only one output driver is active at any time. In the bargraph mode, all drivers from L 00  through the presently decoded zone are active at that time. These modes are summarized in the table on FIG.  19 . 
     L 88  and L 99  come under additional control via an upper flasher  400  which is itself under control of memory locations PR 11 , PR 12 , and the 2-Herz reference from C 16 . There are four upper flashing modes defined by the upper flasher truth table on FIG.  24 . When flashing is required, the output is modulated by C 16  or C 16 . 
     L 11  and L 00  come under additional control via a lower flasher  402  which is itself under control of memory locations PR 13 , PR 14 , and the 2-Herz reference from C 16 . There are four lower flashing modes defined by the lower flasher truth table on FIG.  25 . When flashing is required, the output is modulated by C 16  or C 16 . 
     An upper output  420  (“HI OUT”) is under direct control of the upper two decoded levels and memory locations PR 15  and PR 16 . This output signal is independent from the nature of the output drivers L 88  and L 99  (i.e. LED versus LCD, pointer versus bargraph, flashing mode). There are four “HI OUT” functional modes as described in the HI out truth table on FIG.  26 . 
     A lower output  430  (“LO OUT”) is under direct control of the lower two decoded levels and memory locations PR 17  and PR 18 . This output signal is independent from the nature of the output drivers L 00  and L 11  (i.e., LED versus LCD, pointer versus bargraph, flashing mode). There are four “LO OUT” functional modes as described in the LO out truth table on FIG.  27 . 
     The final discussion describes the transformation of two non-linear resistive transducer characteristics into two new non-linear display characteristics. The first uses an ascending pressure-to-resistance transducer, as indicated on FIG.  28 . The second uses a descending temperature-to-resistance transducer, as indicated on FIG.  29 . Each transducer is located in one leg of the input bridges shown on FIGS. 30 (R P ) and  31  (R T ), respectively. these produce voltage transforms summarized in the tables on FIGS. 32 and 33, respectively. 
     The first step in the process entails characterizing the transducer over its complete operating range, paying particular attention to the regions of primary interest (usually expansion and alarm regions). The second step involves the choice of driving mode, e.g., current source, voltage source, voltage source through an impedance, etc. In these examples, the transducers are characterized resistive transducers (R T ), and the source is voltage through a series resistance. The series resistor (R 1 ) is selected to produce an arbitrary full range voltage compatible with the “XDCR IN” capability while maintaining the desired resolution and accuracy for the task. If the unit is to be made to operate in the ratio mode, the other half of the bridge (R 2  and R 3 ) must be calculated to produce a reference compatible with the range, resolution, and accuracy desired. Note in these examples, the “REF IO” is approximately at full scale relative to the transducer&#39;s maximum output voltage in the bridge shown on FIG. 30, but is at only approximately 70% of the transducer&#39;s maximum output in the bridge shown on FIG.  31 . The 70% truncation is done to improve accuracy and resolution, by allowing the analog-to-digital conversion process to overrange and limit at 255 for values beyond the last programmed threshold. If the same strategy had been applied to FIG. 30, a truncation limit of approximately 80% could have been used with a 4% safety margin. Truncation at the lower voltage end of the scale is also possible, but, in these examples, it would have required means beyond the practical lowest cost solution illustrated. 
     The third step entails calculating the “XDCR IN” voltages at each of the desired output transitions. For example, in the table on FIG. 32, the 100 pounds-persquare-inch-voltage is equal to [(136)/(136+400)]×(5.10)+1.294 volts. The fourth step converts this voltage to a decimal number, and ultimately its equivalent binary number for insertion into non-volatile memory  220 . Again referring to FIG. 32 and 100 pounds per square inch, T 99  is equal to [(1.294)/(1.734)]×(255)=190.3 (round off to 190). 
     The personality of the instrument is then completed by loading all of the functional and calibration instructions into non-volatile memory  220 . The table on FIG. 34 summarizes a completely filled memory for the example defined by FIGS. 28,  30 , and  32 . 
     FIG. 35 illustrates schematically the layout of a single-chip CMOS implementation of the present invention, generally indicated by the reference numeral  500 , which includes all the analog, digital, and memory elements shown on FIGS. 8A,  8 B, and  8 C. Chip  500  has external height and width dimensions of about 0.11 inch by 0.11 inch, with bonding pads measuring about 0.004 inch by 0.004 inch. The manufacturing cost of chip  500  is $1.00 or less, using 1996 costs. 
     In most cases, most, if not all, of the foregoing programming will be completed by the instrument manufacturer; however, programming can also be accomplished at the user level. 
     FIGS. 36-38 illustrate the elements of instrument  230  of FIG. 9 in a physical embodiment. The elements are mounted on a thin, horizontal substrate  600  having a plurality of leads, as at  602 , depending therefrom. The dimensions given on FIGS. 37 and 38 are in inches. It will be understood that a housing (not shown), if any, will contribute slightly to the overall dimensions of the final package. Of course, many other compact arrangements of the elements are possible. As is discussed above, the package is self-contained and, with only connections to power and transducer R x  can selectively provide a variety of outputs on display  232  from virtually any shape input from the transducer. 
     It will thus be seen that the objects set forth above, among those elucidated in, or made apparent from, the preceding description, are efficiently attained and, since certain changes may be made in the above construction without departing from the scope of the invention, it is intended that all matter contained in the above description or shown on the accompanying drawing figures shall be interpreted as illustrative only and not in a limiting sense. 
     It is also to be understood that the following claims are intended to cover all of the generic and specific features of the invention herein described and all statements of the scope of the invention which, as a matter of language, might be said to fall therebetween.