Abstract:
A microwave antenna comprises a single moded metal waveguide tapering inwardly to a cutoff dimension near the distal end thereof. The antenna also comprises a first solid dielectric waveguide mounted coaxially within the distal end portion of the metal waveguide and tapering outwardly toward the inwardly tapering portion of the metal waveguide. The first dielectric waveguide extends beyond the distal end of the metal waveguide in the axial direction. The antenna also comprises a second dielectric waveguide surrounding the first dielectric waveguide beyond the distal end of the metal waveguide and having a dielectric constant lower than the dielectric constant of the first dielectric waveguide. A distal end portion of the first dielectric waveguide tapers inwardly toward the axis thereof, to launch signals propagating toward the distal end of the first dielectric waveguide into the second dielectric waveguide.

Description:
FIELD OF THE INVENTION 
     The present invention relates generally to microwave transitions and antennas of the type that utilize dielectric rods. 
     SUMMARY OF THE INVENTION 
     It is a principal object of the present invention to provide an improved microwave transition for efficiently launching microwave signals from a metallic waveguide into a dielectric waveguide. 
     It is another primary object of the present invention to provide an improved dielectric rod antenna that is capable of producing gains in excess of 20 dB when operated at frequencies of 10 GHz and higher. 
     Another important object of this invention is to provide an improved dielectric rod antenna which produces a pattern having a narrow main lobe and very small side lobes in both the E and H planes. 
     A further object of this invention is to provide an improved dielectric rod antenna which is both small and light weight. 
     Still another object of this invention is to provide such improved microwave transitions and dielectric rod antennas which can be efficiently and economically manufactured. 
     Other objects and advantages of the invention will be apparent from the following detailed description and the accompanying drawings. 
     In accordance with the present invention, the foregoing objectives are realized by providing a microwave transition comprising a single-moded metal waveguide, a dielectric rod mounted coaxially within the distal end portion of the metal waveguide and made of a first dielectric material, a distal portion of the dielectric rod extending beyond the distal end of the metal waveguide, and a second dielectric material surrounding the dielectric rod beyond the distal end of the metal waveguide and having a dielectric constant lower than the dielectric constant of the first dielectric material. An end portion of the dielectric rod tapers inwardly toward the distal end thereof, to launch signals propagating toward the distal end of the dielectric rod into the second dielectric material. 
     The microwave transition of this invention is particularly useful to form a microwave antenna by terminating the second dielectric material at or beyond the distal end of the first dielectric material to radiate the signals launched into the second dielectric material from the dielectric rod, or to receive signals and couple them into the dielectric rod, and then on into the metal waveguide. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is an exploded perspective view of a dielectric rod antenna embodying the present invention; 
     FIG. 2 is an enlarged longitudinal section of the dielectric rod antenna illustrated in FIG. 1; 
     FIG. 3 is a graph of certain parameters relating dielectric rod waveguide to circular metallic dielectric filled waveguide. 
     FIG. 4 is a radiation pattern produced by an exemplary antenna embodying the invention; and 
     FIG. 5 is a longitudinal section of a microwave transition for launching microwave signals for a metallic waveguide into a dielectric waveguide. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     While the invention will be described in connection with certain preferred embodiments, it will be understood that it is not intended to limit the invention to those particular embodiments. On the contrary, it is intended to cover all alternatives, modification and equivalents as may be included within the spirit and scope of the invention as defined by the appended claims. 
     Turning now to the drawings and referring first to FIGS. 1 and 2, there is shown a microwave antenna formed from three components, a metal waveguide 10 including a flared horn 14 on one end, a first dielectric waveguide 11, and a second dielectric waveguide 12. In the transmitting mode, the metal waveguide 10 receives microwave signals from a signal generating source connected to the proximal end of the waveguide, which is the left-hand end as viewed in FIGS. 1 and 2. The metal waveguide 10 preferably has a circular cross section, and is dimensioned so that the fundamental mode of signal propagation is the TE 11  mode, also known as the H 11  mode. The metal waveguide 10 is also preferably dimensioned so that it is single-moded, i.e., modes of higher order than the TE 11  mode are cut off. 
     The distal end portion of the metal waveguide 10 contains the first dielectric waveguide 11, which is in the form of a solid dielectric rod. The dielectric rod 11 preferably has a dielectric constant of less than about 4. One particularly suitable material is Rexolite having an dielectric constant ε of about 2.6. The proximal end portion 11a of the dielectric rod 11 tapers outwardly, and the surrounding portion 10a of the metal waveguide 10 tapers inwardly so as to transfer TE 11  -mode signals to the dielectric rod 11. The inward tapering of the metal waveguide 10 and the outward tapering of the dielectric rod 11 terminate at 13 where the two surfaces meet each other. The minimum diameter of the metal waveguide 10 at 13, where the inward taper is terminated, is preferably less than the cutoff dimension for the TM 01  mode of the dielectric-filled circular waveguide. 
     As an alternative to the transition shown in FIGS. 1 and 2 for coupling energy between the metal waveguide and the dielectric waveguide, a metal waveguide cavity may be coupled at one end to a conventional probe extending into the cavity, and at the other end to the dielectric rod 11. In this case the rod 11 would be terminated within the throat of the horn 14 (i.e., the tapered section at the left-hand end of the rod 11 would be eliminated), and the metal waveguide cavity would have the same transverse cross-sectional size and shape as the rod 11. 
     The distal end portion of the metal waveguide 10 flares outwardly to form the horn 14, for launching signals from the metal waveguide 10 into the first dielectric waveguide 11. The portion of the dielectric rod 11 that is within the horn 14, i.e., between 13 and the distal end of the metal waveguide 10, has a substantially constant diameter. The horn 14 preferably has an exponential taper to remove the metal boundary gradually and ensure that the TE 11  -mode signals are launched into the dielectric rod 11 in the HE 11  mode without any significant radiation from the horn aperture, i.e., the horn aperture is non-radiating at the operating frequency in the absence of the dielectric rod. The horn 14 is terminated at a diameter that is sufficiently large to reduce the evanescent tail of the field of the dielectric waveguide to a level about 40 to 50 dB below the peak value. An exponential horn taper is preferred because the slope is zero at the beginning of the horn, and then changes only gradually at the smaller diameters where the slope is most critical. At the larger diameters the slope is not as critical, and it is at these diameters that the slope of the exponential taper changes most rapidly. A particularly preferred exponential horn taper follows the equation r=exp(ax 2 )-r 0 . 
     Beyond the horn 14, the dielectric rod 11 tapers inwardly at an angle sufficiently small (less than about 5°, preferably less than about 2°) to avoid appreciable radiation from the side surfaces of the rod 11. For a more compact design, the taper of the rod 11 may begin inside the horn 14. As the diameter of the rod 11 diminishes, the field external to the rod expands and is captured by the second dielectric waveguide 12 to form a relatively large antenna aperture. As will be discussed in more detail below, the maximum diameter of the rod 11 is selected to be large enough, for the dielectric constant of the rod material and at the operating frequency, to contain the fields in the rod. The minimum diameter is selected to be small enough to cause most of the energy distribution to be outside the rod 11. The taper between the maximum and minimum diameters, along the length of the rod 11, preferably decreases in slope as the diameter decreases, to minimize radiation from the taper. 
     The physical size of a dielectric waveguide depends on the dielectric constant of the core and the cladding material. The normalized wavenumber, V drwg , of a dielectric rod waveguide is known to be 
     
         V.sub.drwg =k.sub.0 b(ε.sub.1 -ε.sub.2).sup.1/2 
    
     where k 0  =2π/λ 0 , λ 0  is the operating wavelength, and b is the radius of the core. The relative permittivities of the core and cladding material are ε 1  and ε 2 , respectively. The single-mode operating region is 
     
         0&lt;V.sub.drwg &lt;2.405 
    
     However, when V drwg  is too low, the waveguide fields extend very far into the cladding. As a minimum from a practical viewpoint, V drwg  &gt;1. Preferably, V drwg  ≅1.5 so that the field is tightly bounded to the waveguide. When V drwg  &lt;1, a substantial amount of the power is outside the core. Therefore, from practical considerations the single-mode operating range of the dielectric rod waveguide is limited to 
     
         1&lt;V.sub.drwg &lt;2.405 
    
     The single-mode operating range of circular waveguide with perfectly conducting walls is given by 
     
         1,841&lt;V.sub.cwg &lt;2.405 
    
     Here V cwg  =k 0  a(ε 1 ) 1/2   where a is the radius of the metal boundary. The transition from a circular waveguide filled with dielectric having a permittivity ε 1  and operating in the single mode range with radius a, to a dielectric rod waveguide of radius b consisting of the same dielectric material but submerged in the second dielectric material with permittivity ε 2 , also operating in the single-mode regime, is described below. 
     The radius b of a dielectric rod waveguide depends on the ratio between ε 1  and ε 2 . For large ε 1  /ε 2 , the radius is smaller than the radius of the circular waveguide. For small differences in the dielectric, the radius b becomes larger than the largest size allowed in the single-mode regime of the circular metallic waveguide. In this case, the transition from one waveguide to the other without higher-order mode generation is required. The ratio of the normalized wavenumbers is given by ##EQU1## and is plotted in FIG. 3. There is a ratio of dielectric constants when the V drwg  is at the minimum value and the V cwg  is at the maximum value, which defines when ##EQU2## is too small to provide a simple waveguide transition. This occurs at ##EQU3## By reversing the above equation, ##EQU4## the critical ratio ε 1  /ε 2  =1.209 is found. For ratios below this critical value, the radius of the circular metallic waveguide is too large, and therefore overmoded. If the size of the rod is reduced to match the largest allowable size of the circular waveguide, then the operating V drwg  is lower than an acceptable practical value. Returning to FIGS. 1 and 2, the proximal portion of the second dielectric waveguide 12 is formed around the dielectric rod 11, and the distal portion of the waveguide 12 preferably extends beyond the distal end of the rod 11. Alternatively, the dielectric waveguide 12 may terminate at the distal end of the rod 11. This second dielectric waveguide 12 is preferably formed of a foam dielectric so that it has a much smaller dielectric constant than the rod 11, and of course the waveguide 12 also has a larger diameter than the rod 11. The most preferred foam dielectrics are those having dielectric constants below about 4.0. The lower the dielectric constant of this waveguide 12, the larger the mode field distribution and, therefore, the larger the effective antenna aperture and the resultant gain. 
     The presence of the second dielectric waveguide 12 produces a substantial increase in the gain of the antenna, due to the larger mode field of the lower-dielectric-constant waveguide. The magnitude of the gain increase depends upon the diameter of the dielectric and the length of is extension beyond the distal end of the inner rod 11. As illustrated by the broken lines 15a and 15b in FIG. 2, the gain may be further increased by gradually tapering the second waveguide 12 to either increase or decrease its diameter toward the digital end, provided the taper is gradual enough to prevent radiation laterally from the second dielectric. The change in diameter effected by the taper changes the V of the dielectric waveguide, and the maximum gain can be increased by either increasing or decreasing V from a V value at which maximum gain is a minimum. Such tapers are particularly feasible for submillimeter waves because the size of the antenna is so small. 
     The antenna gain can also be increased by the use of multiple concentric sheaths of dielectric material, with each successive sheath having a lower dielectric constant than the adjacent inner sheath. Each sheath is tapered so that it reduces in diameter toward its distal end, and the next outer sheath extends axially beyond the end of its inwardly adjacent sheath. Each time an electromagnetic wave is handed off from one sheath to another, the mode field increases and thus the gain also increases. 
     The field distribution across the aperture of the antenna is approximately described in the rod by the Bessel J 0  function, which is periodic, and in the space surrounding the rod by the Bessel K 0  function, which decreases exponentially with increasing radius. The field distribution described by these functions becomes approximately gaussian when the aperture is sufficiently large, and thus the aperture radiates with a narrow main lobe and low side lobes. The radiation pattern also has rotational symmetry, and thus the first side lobe level is approximately the same in the E and H planes. 
     If desired, either or both of the dielectric waveguides 11 and 12 may be shaped for pattern or polarization control. For example, the inner waveguide 11 may be provided with a slightly elliptical transverse cross-section anywhere on the waveguide; if the induced total phase delay between both polarization senses, due to the geometry, is designed for 90 degrees, the antenna will receive or transmit circular polarization. Alternatively, the cross-sectional shape of the outer dielectric waveguide 12 may be shaped to improve the directivity of the radiation pattern; any resulting relative phase delay between the polarizations can be counteracted by providing a slight deformation in the inner waveguide 11 so that the antenna receives and transmits linearly polarized signals but radiates with a tailored pattern. Although the waveguides 11 and 12 have been illustrated as having circular transverse cross sections, other suitable transverse cross sections are elliptical, oval and rectangular. 
     The normalized wavenumber V in a solid dielectric waveguide is defined by the equation ##EQU5## where d is the diameter of the waveguide, λ 0  is the free space wavelength at the operating frequency, and ε 1  and ε 2  are the dielectric constants of the waveguide material and the material surrounding the waveguide, respectively. 
     For a circular rod, the value of V must be less than 2.4 to cut off modes of higher order than the desired HE 11  mode. In dielectric foam, ε 2  =1.03. Thus, for a Rexolite rod (ε 1  =2.55) surrounded by dielectric foam and operating at a frequency of 28.5 GHz, where λ=1.052 cm., the maximum value of the rod diameter d can be computed as follows: ##EQU6## 
     As a practical matter, the fields outside the rod extend too far when V is less than about 1.5. Thus, for a Rexolite rod in dielectric foam operating at 28.5 GHz, the minimum value of d can be computed as follows: ##EQU7## 
     In order to launch the TE 11  -mode energy into the dielectric waveguide 11, the inside diameter of the metal waveguide 10 is reduced enough to cut off the TM 01  mode when the metal waveguide is filled with the Rexolite dielectric. To achieve this result, the inside diameter of the metal waveguide 10 must be reduced below 0.504 cm at 28.5 GHz. At this diameter, a dielectric material having a relatively high dielectric constant must be used to maintain the value of V above 1.5 and thereby avoid excessive expansion of the field outside the horn. After the signal is in the dielectric waveguide, however, the diameter of the waveguide can be gradually increased. 
     In one example of the invention, an antenna designed for operation at 28.5 GHz had an inner dielectric rod made of Rexolite with a diameter of 0.491 cm and a tapered section 19.3 cm in length and tapering down to a diameter of 0.246 cm. The outer dielectric sheath was made froth an expanded polystyrene foam and the sheath had a diameter of 3.81 cm and a length of 40.64 cm. The dielectric constants of the two dielectrics were 2.55 and 1.03. The V value of the Rexolite red with foam cladding waveguide before the tapered section was 1.8, and at the end of the tapered inner rod the V value was 0.9. The V value of the dielectric sheath with free space cladding waveguide was 2.12. This antenna produced good radiation patterns with a directivity of 25.4 dBi. An exemplary radiation pattern produced by this antenna is shown in FIG. 4 of the drawings. 
     The antenna of this invention is particularly useful in combination with a transmission line in the form of a dielectric waveguide, because signals can be coupled directly between the transmission line and the central inner rod of the antenna. Similarly, the antenna of this invention can be directly coupled to a high-frequency circuit formed from integrated-optics. 
     The transition used in the antenna of FIGS. 1 and 2 for converting the TE 11  mode to the HE 11  mode, and vice versa, is also useful in coupling a dielectric waveguide to a non-dielectric transmission line, such as a metal waveguide. In the transition illustrated in FIG. 5, microwave energy is coupled between a circular metal waveguide 30 and a circular dielectric waveguide 31. The metal waveguide 30 is standard circular waveguide. The dielectric waveguide 31 has a low density foam dielectric cladding 33. Also, the dielectric waveguide 31 has a core 32 made of either a solid dielectric or a foam dielectric slightly higher in density than the foam dielectric cladding 33. A solid dielectric rod 34 within the core 32 extends into the metal waveguide 30, in the same manner as the dielectric rod 11 described above. The rod 34 is gradually tapered toward its distal end before it terminates within the core 32. In the following example, the dielectric waveguide consists of a core of relatively higher density foam than the cladding. The dielectric constant of the cladding foam may be 1.035. The dielectric constant of the core may be 1.12. A dielectric waveguide of this type is desired due to the low loss properties of the foam dielectrics. The ratio of the two dielectric constants 1.082. This ratio is below the critical value of 1.209 and therefore the diameter of the core is larger than the diameter of a single-moded circular metallic waveguide. 
     There is preferably only a small difference between the dielectric constants of the adjoining dielectric materials used in the transition of FIG. 5. For example, the dielectric constants of the inner rod 34, the core 32, and the foam cladding 33 may be 2.55, 1.12 and 1.035, respectively. In a transition using materials having these dielectric constants and designed for operation at 38.5 GHz (λ=1.052 cm), the rod 34 may have a maximum diameter of 0.491 cm tapering down to 0.246 cm at its distal end along a length of 31.4 cm at a taper angle of 0.22°. The core 32 and the cladding 33 may have diameters of 2.296 and 11.483 cm, respectively. The corresponding V values are 1.75 at the larger end of the tapered section of the rod 34, 0.87 at the small end of the tapered section of the rod 34, and 2.0 beyond the end of the rod 34. A particularly preferred dielectric material for the core 32 is isotactic polypropylene, which exhibits low loss characteristics at frequencies such as the 38.5 GHz mentioned above, and higher.