Abstract:
There is provided with an amplifying device including: an input terminal configured to input an input signal; first to ith blocks including first to ith resonators having different first to ith resonance frequencies and first to ith amplifiers that amplify signals which have passed through the first to ith resonators; a divider configured to divide the input signal for the first to ith resonators; a combination section configured to combine the signals which have passed through the first to ith blocks to obtain a combined signal; and an output terminal configured to output the combined signal, wherein a jth (j: an integer between 1 and i−1) block includes a phase adjustment section which provides an output signal of the jth block with a phase difference within a range of {(180±30)+(360×n)} degrees (n: an integer of 0 or greater) from an output signal that passes through a (j+1)th block.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
   This application is based upon and claims the benefit of priority from the prior Japanese Patent Applications No. 2006-112331 filed on Apr. 14, 2006, the entire contents of which are incorporated herein by reference. 
   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to an amplifying device and radio communication circuit. 
   2. Related Art 
   An amplifying device is used to amplify a signal to desired electric field strength. When one amplification device alone cannot amplify a signal to desired power, it is possible to use a method as follow. In this method, a plurality of amplification devices are connected in parallel, then output power of the respective amplification devices is in-phase combined, and thereby desired power is obtained. 
   On the other hand, since an amplification device used in an amplifying device has non-linearity, there is a problem that it outputs non-linear distortion together with the signal. As one scheme for this problem, there is a method of operating the amplification device in an area of low output with good linearity. But this method involves a problem that efficiency deteriorates significantly in the low output area. 
   Therefore, a method called “push-pull” is available to operate the amplifying device with high efficiency whereby non-linear amplification devices are connected in parallel, time of operating each amplification device is changed respectively, and thereby linear output is realized. In addition to this, there is also a method using a linearizer circuit to eliminate distortion outputted from a non-linear amplification device. A general method of using a linearizer circuit is a method using a pre-distortion type linearizer circuit to which a signal having reverse distortion of an amplification device is given beforehand and which gives the sum of the signal having reverse distortion and an input signal to the amplifying device. Furthermore, there is also a feed-forward type linearizer circuit which extracts a distortion signal from an output signal of an amplification device, generates a reverse distortion signal from the extracted distortion signal and subtracts this reverse distortion signal from the output signal of the amplification device to thereby eliminate distortion. 
   However, these methods have such problems that an additional circuit, temperature stabilization circuit or the like is necessary or that the amplifying device configuration is complicated and it requires considerable development time. Furthermore, the output of the amplifying device still includes distortion and it is difficult to eliminate this distortion even when a filter is provided after the amplifying device. 
   SUMMARY OF THE INVENTION 
   According to an aspect of the present invention, there is provided with an amplifying device comprising: 
   an input terminal configured to input an input signal; 
   first to ith blocks including first to ith resonators having respectively different first to ith resonance frequencies (first resonance frequency&lt;second resonance frequency&lt; . . . &lt;ith resonance frequency) and first to ith amplifiers that amplify signals which have passed through the first to ith resonators; 
   a divider configured to divide the input signal for the first to ith resonators; 
   a combination section configured to combine the signals which have passed through the first to ith blocks to obtain a combined signal; and 
   an output terminal configured to output the combined signal, 
   wherein a jth (j: an integer between 1 and i−1) block includes a phase adjustment section which provides an output signal of the jth block with a phase difference within a range of {(180±30)+(360×n)} degrees (n: an integer of 0 or greater) from an output signal that passes through a (j+1)th block. 
   According to an aspect of the present invention, there is provided with an amplifying device comprising: 
   an input terminal configured to input transmission data; 
   first to ith blocks; 
   a signal processing circuit configured to perform transmission processing on the transmission data to generate transmission signal and divide the transmission signal for the first to ith blocks; 
   a power combination section configured to combine signals that have passed through the first to ith blocks to obtain a combined signal; and 
   an output terminal configured to output the combined signal, 
   wherein the first to ith blocks include: 
   first to ith frequency converters configured to convert frequencies of signals that pass through the first to ith blocks; 
   first to ith oscillators configured to give first to ith reference signals having respectively different first to ith frequencies (first frequency&lt;second frequency&lt; . . . &lt;ith frequency) to the first to ith frequency converters; 
   first to ith amplifiers configured to amplify signals that pass through the first to ith blocks; and 
   first to ith resonators configured to have resonance frequencies of the same frequencies as the first to ith frequencies and extract signals depending on the resonance frequencies from the signals amplified by the first to ith amplifiers, 
   wherein jth (j: an integer between 1 and i−1) and (j+1)th oscillators give the jth and (j+1)th reference signals having different phases to the jth and (j+1)th frequency converters to provide the signals that pass through the jth and (j+1)th blocks with phase differences within a range of {(180±30)+(360×n)} degrees (n: an integer of 0 or greater). 
   According to an aspect of the present invention, there is provided with a radio communication circuit comprising: 
   an input terminal configured to input transmission data; 
   first to ith blocks; 
   a signal processing circuit configured to perform transmission processing on the transmission data to generate transmission signal and divide the transmission signal for the first to ith blocks; 
   a power combination section configured to combine signals that have passed through the first to ith blocks to obtain a combined signal; and 
   an output terminal configured to output the combined signal, 
   wherein the first to ith blocks includes: 
   first to ith frequency converters configured to convert frequencies of signals that pass through the first to ith blocks; 
   first to ith oscillators configured to give first to ith reference signals having respectively different first to ith frequencies (first frequency&lt;second frequency&lt; . . . &lt;ith frequency) to the first to ith frequency converters; 
   first to ith amplifiers configured to amplify signals that pass through the first to ith blocks; and 
   first to ith resonators configured to have resonance frequencies of the same frequencies as the first to ith frequencies and extract signals depending on the resonance frequencies from the signals amplified by the first to ith amplifiers, 
   wherein a jth (j: an integer between 1 and i−1) block further includes a phase adjustment section which provides an output signal of the jth block with a phase difference within a range of {(180±30)+(360×n)} degrees (n: an integer of 0 or greater) from an output signal that passes through a (j+1)th block. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a block diagram showing an embodiment of an amplifying device of the present invention; 
       FIG. 2  illustrates principles of power combination; 
       FIG. 3  illustrates a frequency response characteristic when opposite phases are combined; 
       FIG. 4  illustrates a frequency response characteristic when in-phase combination is performed; 
       FIG. 5  illustrates an output frequency response characteristic when two sine wave signals of the same level are input to a non-linear amplification device; 
       FIG. 6  illustrates an output frequency response characteristic when two sine wave signals of different levels are input to a non-linear amplification device; 
       FIG. 7  illustrates an output frequency response characteristic when a modulated signal is inputted to a non-linear amplification device; 
       FIG. 8  illustrates an output frequency response characteristic at outputs of the amplification devices when a modulated signal is inputted to the circuit in  FIG. 1  (i=4); 
       FIG. 9  illustrates an output frequency response characteristic when a modulated signal is inputted to the circuit in  FIG. 1 ; 
       FIG. 10  illustrates an output frequency response characteristic of the circuit in  FIG. 1  (i=6); 
       FIG. 11  illustrates an output frequency response characteristic of the circuit in  FIG. 1  (i=4); 
       FIG. 12  is a circuit diagram showing a specific configuration example of the amplifier; 
       FIG. 13  illustrates an I-V curve in a field effect transistor; 
       FIG. 14  is a block diagram showing an embodiment of the amplifying device of the present invention; 
       FIG. 15  is a block diagram showing another embodiment of the amplifying device of the present invention; 
       FIG. 16  is a block diagram showing a further embodiment of the amplifying device of the present invention; 
       FIG. 17  is a block diagram showing a still further embodiment of the amplifying device of the present invention; 
       FIG. 18  illustrates a modulated signal when GFSK is used; 
       FIG. 19  is a block diagram showing a still further embodiment of the amplifying device of the present invention; 
       FIG. 20  is a circuit diagram showing a specific configuration example of the amplifier; 
       FIG. 21  illustrates an output frequency response characteristic of the embodiment shown in  FIG. 19 ; 
       FIG. 22  is a block diagram showing a still further embodiment of the amplifying device of the present invention; 
       FIG. 23  is a block diagram showing a still further embodiment of the amplifying device of the present invention; 
       FIG. 24  illustrates a configuration example when the amplifying device according to an embodiment of the present invention is applied to a transmission section of a radio communication device; 
       FIG. 25  illustrates a configuration example when the amplifying device according to an embodiment of the present invention is applied to a reception section of the radio communication device; 
       FIG. 26  is a block diagram showing an embodiment of a radio communication circuit of the present invention; 
       FIG. 27  is a block diagram showing another embodiment of the radio communication circuit of the present invention; 
       FIG. 28  illustrates a configuration example of a power divider; 
       FIG. 29  illustrates another configuration example of the power divider; and 
       FIG. 30  illustrates a configuration example of a branching filter. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1  shows a first embodiment of an amplifying device according to the present invention. 
   This amplifying device is provided with an input terminal  98 , a power divider  105 , blocks BL( 1 ) to BL(i) (i: an integer of 2 or greater), a power combiner (power combination section)  106  and an output terminal  99 . The blocks BL( 1 ) to BL(i) are connected in parallel to the power divider  105  on the input side and connected in parallel to the power combiner  106  on the output side. 
   The block BL(X) (X=1 to i) includes a resonator  101 (X), a variable phase section (phase adjustment section)  102 (X), a variable amplitude section (amplitude adjustment section)  103 (X), an amplifier  104 (X) and a resonator  1001 (X). Examples of the resonator  101 (X) and resonator  1001 (X) include a cavity type, dielectric resonator type, transmission line type having transmission line of m (m: an integer of 1 or greater) times the half-wavelength and concentrated constant type using an inductor and capacitor or the like. A superconductor may also be used as a conductive part of the resonator  101 (X) and resonator  1001 (X). In this case, a signal of steeper shape can be extracted. A resonator using a superconductor is especially effective for use in a narrow-band radio system. The power divider  105  divides an input signal inputted from the input terminal  98  for the blocks BL( 1 ) to BL(i). The signals which have passed through the blocks BL( 1 ) to BL(i) are combined in power by the power combiner  106  and the combined signal is outputted from the output terminal  99 . 
   The resonators  101 ( 1 ) to  101 ( i ) in the blocks BL( 1 ) to BL(i) have different resonance frequencies f 1 , f 2 , . . . , fi. Assuming that there is a relationship of f 1 &lt;f 2 &lt; . . . &lt;fi, an interval between resonance frequencies may or may not be constant. In  FIG. 1 , blocks are arranged in ascending order of resonance frequency, but the block arrangement order is arbitrary. The resonators  101 ( 1 ) to  101 ( i ) are connected to the amplifiers  104 ( 1 ) to  104 ( i ) through the variable phase sections  102 ( 1 ) to  102 ( i ) and variable amplitude sections  103 ( 1 ) to  103 ( i ). The order of connection between the variable phase section, variable amplitude section and amplifier is arbitrary and, for example, the positions of the variable phase section and variable amplitude section may be switched round. The outputs of the amplifiers  104 ( 1 ) to  104 ( i ) are connected to the resonators  1001 ( 1 ) to  1001 ( i ) having the same resonance frequencies f 1 , f 2 , . . . , fi as those of the resonators  101 ( 1 ) to  101 ( i ). 
   Here, the operations of the variable phase sections  102 ( 1 ) to  102 ( i ) and variable amplitude sections  103 ( 1 ) to  103 ( i ) are set such that when the signals (output signals of resonators  1001 ( 1 ) to  1001 ( i )) which have passed through the blocks BL( 1 ) to BL(i) are combined in power between neighboring resonance frequency blocks at the power combiner  106  (that is, at a power combination point), combined neighboring signals have a phase difference within a range of (180±30)+360×n degrees (n: an integer of 0 or greater) (substantially opposite phase) and the same amplitude. This allows the signals which have passed through the blocks BL( 1 ) to BL(i) to be combined appropriately. Here, the operations of the variable phase sections  102 ( 1 ) to  102 ( i ) and variable amplitude sections  103 ( 1 ) to  103 ( i ) are set through a control signal from outside. 
   In operation, a signal inputted from the input terminal  98  is divided for the resonators  101 ( 1 ) to  101 ( i ) by the power divider  105 . Signals corresponding to the resonance frequencies f 1 , f 2 , . . . , fi of the resonators  101 ( 1 ) to  101 ( i ) are extracted at the resonators  101 ( 1 ) to  101 ( i ) and the extracted signals are given to the amplifiers  104 ( 1 ) to  104 ( i ) through the variable phase sections  102 ( 1 ) to  102 ( i ) and variable amplitude sections  103 ( 1 ) to  103 ( i ). The signals given to the amplifiers  104 ( 1 ) to  104 ( i ) are amplified according to gains of the amplifiers  104 ( 1 ) to  104 ( i ). Intermodulation distortion is generated during amplification, but intermodulation distortion is suppressed to a small level for reasons which will be described later. The signals amplified at the amplifiers  104 ( 1 ) to  104 ( i ) are given to the resonators  1001 ( 1 ) to  1001 ( i ) and the resonators  1001 ( 1 ) to  1001 ( i ) remove intermodulation distortion generated at the amplifiers  104 ( 1 ) to  104 ( i ) as well as distortion generated due to active elements at the variable phase sections  102 ( 1 ) to  102 ( i ) and variable amplitude sections  103 ( 1 ) to  103 ( i ). The signals rid of distortion at the resonators  1001 ( 1 ) to  1001 ( i ) are given to the power combiner  106 , combined in power at the power combiner  106  and outputted as a combined signal from the output terminal  99 .  FIG. 9  shows an example of frequency response when a certain modulated signal is inputted to the amplifying device in  FIG. 1 . 
   Hereinafter, the amplifying device in  FIG. 1  will be explained more specifically. 
   First, power combination at the power combiner  106  will be explained in detail using  FIG. 2  to  FIG. 4 . 
     FIG. 2  is a circuit diagram illustrating the principles of power combination. 
   A resonator  205  having a resonance frequency f 1  and a resonator  206  having a resonance frequency f 2  are connected in parallel. Reference numeral  201  denotes input terminals,  202  denotes output terminals,  203  denotes a power divider,  204  denotes a power combiner,  207   a  denotes a coupling circuit which couples the resonator  205  with the power divider  203 ,  207   b  denotes a coupling circuit which couples the resonator  206  with the power divider  203 ,  208   a  denotes a coupling circuit which couples the resonator  205  with the power combiner  204  and  208   b  denotes a coupling circuit which couples the resonator  206  with the power combiner  204 . 
     FIG. 3  shows a frequency response when coupling M 2  between the resonator  206  and coupling circuit  208   b  is reverse phase coupling (phase is inverted by 180 degrees) and a certain signal is inputted to the input terminals  201 . Reference numeral  302   a  denotes a signal extracted by the resonator  205 ,  302   b  denotes a signal extracted by the resonator  206  and  301   a  denotes a signal (combined signal) outputted from the output terminal  202 .  FIG. 4  shows a frequency response at the output terminals  202  when coupling M 2  between the resonator  206  and coupling circuit  208   b  is in-phase coupling (phase remains unchanged) and the above described certain signal is inputted to the input terminal  201 . Reference numeral  302   a  denotes a signal extracted by the resonator  205 ,  302   b  denotes a signal extracted by the resonator  206  and  301   b  denotes a signal outputted from the output terminals  202 . However, suppose coupling m 1  (1) between the resonator  205  and coupling circuit  207   a , coupling m 1  (2) between the resonator  205  and coupling circuit  208   a  and coupling M 2  between the resonator  206  and coupling circuit  207   b  are in-phase coupling. 
   When coupling M 2  between the resonator  206  and coupling circuit  208   b  is in-phase coupling, that is, when two signals to be combined are in phase, the signal amplitude near a center frequency in a target band is lowered as shown in  FIG. 4  and a desired signal cannot be obtained. This is for the following reason: A conventional amplifying device which is made up of amplification devices simply connected in parallel performs power combination in phase, but with the amplifying device in  FIG. 1 , phases of signals before and after a resonance frequency as a boundary are inverted at the resonators  101 ( 1 ) to  101 ( i ) respectively. For this reason, when power combination is performed in phase, a signal with the amplitude in the vicinity of a center frequency lowered in a target band is obtained as shown in  FIG. 4 . On the contrary, when coupling M 2  between the resonator  206  and coupling circuit  208   b  is reverse phase coupling, phases of two signals having mutually neighboring resonance frequencies become opposite phases before combination, and therefore it is possible to obtain a desired signal as shown in  FIG. 3 . A desired signal can be obtained even when the two signals to be combined are not completely in opposite phase, if they are substantially in opposite phase, that is, they have a phase difference within a range of (180±30)+360×n degrees (n: an integer of 0 or greater). 
   Based on the above described principles, to obtain a desired output signal, the amplifying device shown in  FIG. 1  sets the operation of the variable phase sections so that signals which pass through blocks of neighboring resonance frequencies are substantially in opposite phase. Furthermore, when power combination is performed, amplitudes of the signals to be combined need to be the same at a power combination point, and therefore the operation of the variable amplitude sections between the blocks of neighboring resonance frequencies is set so as to satisfy this condition. If amplitudes of two signals to be combined can be made the same at the power combination point, the variable amplitude sections may be omitted. 
   Next, intermodulation distortion generated at the amplifiers  104 ( 1 ) to  104 ( i ) will be explained. 
     FIG. 5  shows an example of an output spectrum when two sine wave signals having the same signal level and different frequencies are inputted to a non-linear amplification device. Solid lines in the figure show output signals when the two inputted sine wave signals are amplified, dotted lines show third-order intermodulation distortion resulting from the two sine wave signals. As is clear from this figure, third-order intermodulation distortion appears over a range three times a signal band Δf (here, frequency difference between the two sine wave signals).  FIG. 6  shows an output spectrum when two sine wave signals of different signal levels (signal strength of one sine wave signal is the same as that in the case of  FIG. 5  and signal strength of the other sine wave signal is smaller than this) are inputted to a non-linear amplification device. It is understandable that third-order intermodulation distortion has been considerably reduced compared to the case in  FIG. 5 . Furthermore, though not shown, the same applies to fifth or higher-order intermodulation distortion. 
     FIG. 7  shows an example of an output spectrum when a modulated signal having a certain band (having certain signal strength over an entire band) is inputted to a non-linear amplification device. As in the case of  FIG. 5 , intermodulation distortion appears over a frequency range three times the band of the input signal. On the other hand,  FIG. 8  shows a spectrum in a case where the outputs from the amplifiers  104 ( 1 ) to  104 ( 4 ) are superimposed on one another when the above described certain modulated signal is inputted from the input terminal  98  assuming i=4 in  FIG. 1 . 
   The signals which have passed through the resonators  101 ( 1 ) to  101 ( 4 ) correspond to those obtained by dividing the above described certain modulated signal into four bands. Each signal has a shape similar to a sine waveform (that is, the signal strength of each signal is highest at the center and decreases as it goes away from the center in  FIG. 8 ) and also has a narrower band than the band of the above described certain modulated signal. In this way, each signal has a shape similar to a sine waveform which is hard to be distorted and has a narrow band, and therefore the signal level of intermodulation distortion generated from each signal is also reduced and the signal band of each intermodulation distortion is also narrowed. Therefore, it is possible to obtain an output signal with less distortion by combining the outputs of the amplifiers  104 ( 1 ) to  104 ( 4 ) as is also understandable from a comparison between  FIG. 7  and  FIG. 8 . However, in  FIG. 1 , the resonators  1001 ( 1 ) to  1001 ( i ) are disposed after the amplifiers  104 ( 1 ) to  104 ( i ), and therefore these resonators  1001 ( 1 ) to  1001 ( i ) eliminate intermodulation distortion generated at the amplifiers  104 ( 1 ) to  104 ( i ) as well as distortion resulting from active elements at the variable phase sections  102 ( 1 ) to  102 ( i ) and variable amplitude sections  103 ( 1 ) to  103 ( i ). Therefore, it is possible to obtain an output signal (combined signal) with considerably reduced distortion according to this embodiment. 
     FIG. 10  illustrates that the amplifying device in  FIG. 1  allows a signal in a target band to pass and prevents signals out of the target band from passing. More specifically,  FIG. 10  shows an example of frequency response assuming i=6 in the circuit of  FIG. 1  and assuming that the gains of the amplifiers  104 ( 1 ) to  104 ( 6 ) are 0 dB (frequency responses at the input terminal  98  and output terminal  99 ). 
   Here, a flat signal (signal having constant signal strength over the entire frequency band) is used as the input signal to the input terminal  98 . Furthermore, the resonance frequencies f 1  to f 6  of the resonators  101 ( 1 ) to  101 ( 6 ),  1001 ( 1 ) to  1001 ( 6 ) are assumed to be f 1 =1.9812 GHz, f 2 =1.988 GHz, f 3 =1.9953 GHz, f 4 =2.0047 GHz, f 5 =2.012 GHz, f 6 =2.0188 GHz. Furthermore, a coupling Q value (external coupling Q value) for coupling the resonators  101 ( 1 ) to  101 ( 6 ) with an external circuit (coupling circuit for coupling the resonator with the power divider and coupling circuit for coupling the resonator with the variable phase section) is set to Qe=400. Likewise, a coupling Q value for coupling the resonators  1001 ( 1 ) to  1001 ( 6 ) with an external circuit (coupling circuit for coupling the resonator with the amplifier and coupling circuit for coupling the resonator with the power combiner) is set to Qe=400. Here, the degree of coupling of the resonator will be defined. Assuming that the coupling Q value on the input side of the resonator is Q in  and the coupling Q value on the output side of the resonator is Q out , the degree of coupling of the resonator is expressed by 1/(1/Q in +1/Q out ). For example, the degree of coupling of the resonator  101 ( 1 ) is 1/(1/400+1/400)=200. 
     FIG. 10  shows a graph (reflection characteristic graph) G 11  (frequency response at the input terminal  98 ) described in a coordinate system where the horizontal axis shows a frequency and the vertical axis shows an S 11  parameter (=reflected signal voltage/input signal voltage).  FIG. 10  still shows a graph (passage characteristic graph) G 21  (frequency response at the output terminal  99 ) described in a coordinate system where the horizontal axis shows a frequency and the vertical axis shows an S 21  parameter (=output signal voltage/input signal voltage). As is understandable from these characteristic graphs G 11 , G 21 , according to this embodiment, signals in a target band are allowed to pass and passage of most signals in non-target bands is rejected (reflected by an input terminal T 1 ). Here, an example where the gains of the amplifiers  104 ( 1 ) to  104 ( 6 ) are set to 0 dB has been shown, but by increasing the gains of the amplifiers  104 ( 1 ) to  104 ( 6 ), the characteristic graphs G 11 , G 21  are obtained in the form shifted up and down accordingly. Thus, according to this embodiment, it is understandable that a signal in a target band is amplified and outputted with low distortion, while passage of signals in non-target bands is blocked. 
     FIG. 11  shows an example of frequency response (frequency responses at the input terminal  98  and output terminal  99 ) when four resonators  101 ( 1 ) to  101 ( 4 ) and four resonators  1001 ( 1 ) to  1001 ( 4 ) are disposed and coupling Q values Qe of resonators differ from one block to another (the degree of coupling of resonators differs from one block to another). 
   However, as in the case of  FIG. 10 , suppose the gains of the amplifiers  104 ( 1 ) to  104 ( 4 ) are set to 0 dB, resonance frequency f 1  of the resonators  101 ( 1 ),  1001 ( 1 ) is 1.988 GHz, resonance frequency f 2  of the resonators  101 ( 2 ),  1001 ( 2 ) is 1.9958 GHz, resonance frequency f 3  of the resonators  101 ( 3 ),  1001 ( 3 ) is 2.0042 GHz and resonance frequency f 4  of the resonators  101 ( 4 ),  1001 ( 4 ) is 2.012 GHz. Furthermore, the coupling Q values of the resonators  101 ( 1 ),  1001 ( 1 ),  101 ( 4 ),  1001 ( 4 ) are supposed to be Qe 1 , Qe 4 =500, the coupling Q values of the resonators  101 ( 2 ),  1001 ( 2 ),  101 ( 3 ),  1001 ( 3 ) are supposed to be Qe 2 , Qe 3 =400. That is, the coupling Q values of the resonators at both ends of the target band are increased (the degree of coupling is increased). In this way, by increasing the coupling Q values of the resonators at both ends of the target band, it is possible to increase the amount of attenuation outside the target band as is also understandable from a graph G 11   a  of the S 11  parameter and graph G 21   a  of the S 21  parameter. 
     FIG. 12  shows an example of the specific configuration of the amplifiers  104 ( 1 ) to  104 ( i ). 
   An example using a field effect transistor  401  as the amplification device is shown. The source of the field effect transistor  401  is grounded. An input matching circuit  402  is connected to the gate of the field effect transistor  401  and an output matching circuit  403  is connected to the drain of the field effect transistor  401 . One end of a bias circuit  404   a  is connected to the gate of the field effect transistor  401  and the other end is connected to a gate terminal  408 . One end of a bias circuit  404   b  is connected to the drain of the field effect transistor  401  and the other end is connected to a drain terminal  407 . An example using a resistor as the bias circuit  404   a  and using an inductor as the bias circuit  404   b  is shown here. Reference numeral  405  denotes an input terminal and  406  denotes an output terminal. A negative voltage is applied to the gate terminal  408 , a positive voltage is applied to the drain terminal  407 , a signal is inputted from the input terminal  405  and an output signal is obtained from the output terminal  406 . 
     FIG. 13  shows an I-V (drain current-drain voltage) curve of the field effect transistor  401 . As is understandable from this I-V curve, it is possible to make a current of the transistor variable by changing the gate voltage. Furthermore, as shown by an arrow in the figure, it is possible to change power consumption by changing the operating point of the transistor. When linearity is required, the transistor is operated as class A and when efficiency is required, the transistor is operated as class B. When an intermediate characteristic is required, the transistor is operated as class AB. This allows power consumption of the amplification device itself to be controlled. Furthermore, by setting the operating point of the transistor at the amplifiers  104 ( 1 ) to  104 ( i ) according to an energy density of signals inputted to the amplifiers  104 ( 1 ) to  104 ( i ), it is possible to realize high operation efficiency. This method is effective especially when the modulation scheme changes with time. For example, when an energy distribution on the frequency axis changes such that QPSK (Quadrature Phase Shift Keying) modulation is used for a certain time period and GMSK (Gaussian filtered Minimum Shift Keying) modulation is used for another time period, it is possible to realize amplification with less efficiency deterioration under each modulation scheme by changing the operating point of the transistor. Furthermore, when a bipolar type device is used, similar effects can be obtained by changing the base voltage. 
     FIG. 14  shows a second embodiment of the amplifying device according to the present invention. 
   Using delay circuits (delayers)  107 ( 1 ) to  107 ( 4 ) as the variable phase sections in  FIG. 1 , this amplifying device is designed so that signals passing through blocks of neighboring resonance frequencies have a phase difference within a range of (180±30)+360×n degrees (n: an integer of 0 or greater) at a power combination point. When, for example, a phase delay of  1800  occurs in each of resonators  101 ( 1 ) to  101 ( 4 ), the delay circuits  107 ( 2 ),  107 ( 4 ) generate a phase delay of 180° and the delay circuits  107 ( 1 ),  107 ( 3 ) generate a phase delay of 0°. In this way, it is possible to obtain an output signal with less distortion by using delay circuits not including any active elements (transistors). In  FIG. 1 , the resonators  1001 ( 1 ) to  1001 ( i ) are disposed on the output side to realize smaller distortion but no resonator is disposed on the output side in this embodiment. When the distortion specification is relatively not stringent, it is possible to obtain an output signal at a degree that does not cause trouble with communications even with the resonator on the output side omitted. 
     FIG. 15  shows a low pass filter  108  disposed after the power combiner  106  as a measure against harmonics generated at the amplifiers  104 ( 1 ) to  104 ( 4 ) in  FIG. 14 . By inputting a combined signal obtained from the power combiner  106  to the low pass filter  108 , it is possible to cut harmonics generated at the respective amplifiers. Effects similar to those of the low pass filter can also be obtained by replacing the low pass filter by a band pass filter. It is of course possible to add a low pass filter or band pass filter to the configuration in  FIG. 1 . 
     FIG. 16  shows a third embodiment of the amplifying device according to the present invention. 
   An example where the number of amplification devices varies among the amplifiers  104 ( 1 ) to  104 ( 5 ) is shown. The amplifier  104 ( 2 ) includes two cascade connected amplification devices  1004   a ,  1004   b . The amplifier  104 ( 3 ) includes three cascade connected amplification devices  1004   c ,  1004   d ,  1004   e . The amplifier  104 ( 4 ) includes two cascade connected amplification devices  1004   f ,  1004   g . The amplifier  104 ( 1 ) and amplifier  104 ( 5 ) each include only a single amplification device. Generally, a modulated signal (communication signal) is occasionally a signal with energy uniformly distributed over a band as in  FIG. 7  and is occasionally a signal having large energy near the center frequency and small energy at an end of the band. Thus, the numbers of amplification devices in the respective amplifiers  104 ( 1 ) to  104 ( 5 ) are determined according to the respective signal levels. This allows amplification devices in each block to be operated at optimum efficiency. This also makes it possible to reduce variable widths of variable amplitude sections  103 ( 1 ) to  103 ( 5 ). 
     FIG. 17  shows an example where amplification devices (transistors) of different sizes (output power levels) are used at their respective amplifiers. More specifically, the final stage amplification devices differ in size from one amplifier to another. The amplifier  104 ( 1 ) has two amplification devices  1009   a ,  1009   b  of the same size. The amplifier  104 ( 2 ) has two amplification devices  1100   a ,  1100   b  of different sizes and the size of the posterior amplification device  1100   b  is greater than the sizes of the anterior amplification device  1100   a  and the amplification device  1009   b  of the amplifier  104 ( 1 ). The amplifier  104 ( 3 ) has two amplification devices  1101   a ,  1101   b  of the same size and the size of amplification device  1101   b  is smaller than the size of the amplification device  1100   b  of the amplifier  104 ( 2 ). When the modulated signal (communication signal) is divided into a plurality of bands on the frequency axis by resonators  101 ( 1 ) to  101 ( 3 ) and the divided signals are amplified, large amplification energy is necessary for a wide band signal. Thus, by disposing an amplification device of appropriate size according to the size of the band of a signal or the size of a signal level in an amplifier, it is possible to operate the amplification device efficiently. 
   As explained using  FIG. 16  and  FIG. 17  above, it is possible to operate the amplifier efficiently by changing the number of amplification devices and the size thereof, for each amplifying device. This will be explained in further detail as follows. 
   One of cellular phone systems used worldwide and centered in Europe at present is GSM (Global System for Mobile Communications). In GMSK (Gaussian filtered Minimum Shift Keying) which is the modulation scheme used by this GSM and GFSK (Gaussian filtered Frequency Shift Keying) used by Bluetooth (trademark), side lobes (band limiting) is cut using a Gaussian filter for narrow band transmission. For this reason, a distribution of modulated signal has a high energy density at the center frequency part.  FIG. 18  shows an example of the modulated signal using GFSK. The horizontal axis shows normalized frequency, and zero of the horizontal axis is a center frequency. Signals with BT=0.3, BT=0.5, BT=1.0 are shown. “BT” (Bandwidth Time) denotes a normalized bandwidth of a Gaussian filter and means that the bandwidth broadens as the BT reduces. As the BT decreases between BT=1.0 and 0.5, the energy density increases in the center frequency part, and in BT&lt;0.5, its relationship turns over. When a signal with BT=0.3 is used as an input signal to an amplifying device and divided on the frequency axis, there is a big difference in the output power level between the amplifier of a band near the center frequency and the amplifier of a band near the end. For this reason, if amplification is performed by amplification devices of the same size in each band, the efficiency of the amplifier in the band at the end deteriorates. Thus, it is possible to increase the overall amplification efficiency by performing amplification using amplification devices of different output power levels according to the energy distribution of a modulated signal. For example, when four resonators having the same coupling Q are used on the input side, the amplifiers in the blocks corresponding to the bands at both ends of the four split bands use amplification devices having approximately 80% of the output power level of the amplification devices used in amplifiers in other blocks. 
     FIG. 19  shows a fourth embodiment of the amplifying device according to the present invention. 
   Amplifiers  104 ( 1 ) to  104 ( 4 ) include switching sections which switch ON/OFF (operating/non-operating state) the own operations. An external control apparatus  109  issues a control signal and in response to the control signal, amplifiers  104 ( 1 ) to  104 ( 4 ) switch the own operating states using the switching sections.  FIG. 20  shows a configuration example of an amplifying device whose operating state can be changed. A switch  409  is provided between a drain terminal  407  and a bias circuit  404   b  as a switching section. A voltage (external control signal) given from the external control apparatus  109  to a control terminal  410  causes the switch  409  to switch ON/OFF a power supply from the drain terminal  407 , and it is possible to thereby change the operating state of a field effect transistor  401 . As other configuration, there is also another configuration in which the field effect transistor  401  is switched OFF by changing a negative voltage on the gate side to a value at which the field effect transistor  401  is placed in a pinch-off state.  FIG. 21  shows a simulation result (passage characteristic) conducted while changing the number of amplifiers to be operated using the amplifying device shown in  FIG. 19 . The horizontal axis shows a frequency and the vertical axis shows an S 21  parameter. A simulation result when all the four amplifiers  104 ( 1 ) to  104 ( 4 ) are operated is shown with a graph G 111 , a simulation result when only the three amplifiers  104 ( 2 ) to  104 ( 4 ) are operated is shown with a graph G 112  and a simulation result when only the two amplifiers  104 ( 3 ),  104 ( 4 ) are operated is shown with a graph G 113 . As is understandable from the graphs G 111  to G 113 , it is possible to obtain a desired output signal by operating only necessary amplifiers in accordance with a signal band to be transmitted. That is, according to this configuration, it is also possible to maintain the operation at high efficiency for signals of different bandwidths. 
     FIG. 22  shows a fifth embodiment of the amplifying device according to the present invention. 
   This amplifying device includes harmonics processing circuits  112 ( 1 ) to  112 ( 4 ) on the output side of amplifiers  104 ( 1 ) to  104 ( 4 ). Intermodulation distortion at the amplifiers  104 ( 1 ) to  104 ( 4 ) is reduced for the aforementioned reason, but signal energy clipped by the amplifiers  104 ( 1 ) to  104 ( 4 ) is outputted as harmonics. Thus, by providing the harmonics processing circuits  112 ( 1 ) to  112 ( 4 ) on the output side of the amplifiers  104 ( 1 ) to  104 ( 4 ), it is possible to increase the efficiency of the amplifiers  104 ( 1 ) to  104 ( 4 ). As an example of the harmonics processing circuit, it is possible to realize a circuit which performs a class F operation by shorting even harmonics and leaving open odd harmonics, and ideally it is possible to realize efficiency of 100%. By connecting such a circuit to the output side of each amplifier, it is possible to operate each amplifier with high efficiency. 
     FIG. 23  shows a sixth embodiment of the amplifying device according to the present invention. 
   In this amplifying device, each resonator includes delay means. Resonators of neighboring resonance frequencies are a (0.5+K) wavelength (K: an integer of 0 or greater) resonator and an L wavelength (L: an integer of 1 or greater) resonator.  FIG. 23  shows an example where K=0, L=1. Here, a resonator  113 ( 1 ) is a half-wavelength resonator having a transmission line having the half-wavelength of its resonance frequency f 1 , a resonator  114 ( 1 ) is a 1-wavelength resonator having a transmission line having the 1-wavelength of its resonance frequency f 2 , a resonator  113 ( 2 ) is a half-wavelength resonator having a transmission line having the half-wavelength of its resonance frequency f 3  and a resonator  114 ( 2 ) is a 1-wavelength resonator having a transmission line having the 1-wavelength of its resonance frequency f 4 . When neighboring resonators satisfy the above described conditions related to K and L, each resonator may has an arbitrary transmission path length. For example, the resonator  113 ( 1 ) may be a 0.5-wavelength resonator. the resonator  114 ( 1 ) may be a 1-wavelength resonator. Or, the resonator  113 ( 1 ) may be a 1.5-wavelength resonator and the resonator  114 ( 2 ) may be a 2-wavelength resonator. 
   Using  FIG. 24  and  FIG. 25 , an example where the amplifying device explained so far ( FIG. 1 ,  FIG. 14 ,  FIG. 15 ,  FIG. 16 ,  FIG. 17 ,  FIG. 19 ,  FIG. 22 ,  FIG. 23 ) is incorporated in a radio communication device will be explained. 
     FIG. 24  schematically shows a configuration example of a transmission section in a radio communication device. Data to be transmitted  500  is inputted to a signal processing circuit  501 , subjected to transmission processing such as digital/analog conversion, coding and modulation, and thereby a transmission signal having a baseband or intermediate frequency (IF) band is generated. The transmission signal generated at the signal processing circuit  501  is inputted to a frequency converter (mixer)  502 , multiplied by a local signal from a local signal generator  503 , and thereby converted to a signal of a radio frequency (RF) band, that is, up-converted. The RF signal outputted from the mixer  502  is amplified by a power amplifier (PA)  504  as an amplifying device according to this proposal and then inputted to a band limiting filter (transmission filter)  505 . The RF signal amplified by the power amplifier  504  is limited in band at this filter  505 , deprived of an unnecessary frequency component and then radiated from an antenna  506  into space as a radio wave. If an unnecessary frequency component can be eliminated in the amplifier  504  according to this proposal, the band limiting filter  505  may not be provided. 
     FIG. 25  schematically shows a configuration example of a reception section in a radio communication device. A signal received through an antenna  506  is inputted to a band limiting filter (reception filter)  508 , limited in band at this reception filter  508 , deprived of an unnecessary frequency component and then inputted to a low noise amplifier (LNA)  507  as an amplifying device according to this proposal. The signal amplified by the low noise amplifier  507  is inputted to a mixer  502 , multiplied by a local signal from a local signal generator  503  and thereby converted to a baseband or intermediate frequency. The signal converted to a low frequency through this conversion is inputted to a signal processing circuit  501 , subjected to demodulation processing, and thereby received data  509  is outputted. If an unnecessary frequency component can be eliminated at the low noise amplifier (LNA)  507 , the band limiting filter  508  may not be provided. 
     FIG. 26  shows an embodiment of a radio communication circuit according to the present invention. 
   This radio communication circuit is provided with an input terminal  91 , a signal processing circuit  501 , blocks BL( 101 ) to BL( 103 ), a power combiner  106  and an output terminal  92 . The blocks BL( 101 ) to BL( 103 ) have mixers  502 ( 1 ) to  502 ( 3 ), oscillators  503 ( 1 ) to  503 ( 3 ), amplifiers  104 ( 1 ) to  104 ( 3 ) and resonators  1001 ( 1 ) to  1001 ( 3 ). The blocks BL( 101 ) to BL( 103 ) are connected to the signal processing circuit  501  in parallel on the input side. The blocks BL( 101 ) to BL( 103 ) are connected to the power combiner  106  in parallel on the output side. 
   The signal processing circuit  501  performs transmission processing such as digital/analog conversion, coding and modulation to the data inputted from the input terminal  91 , and thereby generates a transmission signal of a baseband or intermediate frequency (IF) band and divides (distributes) the transmission signal generated for the blocks BL( 101 ) to BL( 103 ). For example, the signals given to the respective blocks ( 101 ) to BL( 103 ) are the same. 
   The mixers  502 ( 1 ) to  502 ( 3 ) in the blocks BL ( 101 ) to BL ( 103 ) up-convert the signals given from the signal processing circuit  501  using a reference signal from the oscillators  503 ( 1 ) to  503 ( 3 ) to a radio frequency (RF) band. The signals of the respective radio frequencies are amplified by the amplifiers  104 ( 1 ) to  104 ( 3 ) and then given to the resonator  1001 ( 1 ) to resonator  1001 ( 3 ). Here, the oscillating frequencies of the oscillators  503 ( 1 ) to  503 ( 3 ) are the same as the resonance frequencies of the resonators  1001 ( 1 ) to  1001 ( 3 ). That is, all the oscillating frequencies of the oscillators  503 ( 1 ) to  503 ( 3 ) are different. Furthermore, due to the placement of the resonators  1001 ( 1 ) to  1001 ( 3 ), the signals which have passed through the respective blocks of neighboring resonance frequencies need to have a phase difference within a range of (180±30)+360×n degrees (n: an integer of 0 or greater) of the same amplitude at a power combination point of the power combiner  106 . For this reason, the oscillators  503 ( 1 ) to  503 ( 3 ) are made to oscillate with the amplitude and phase that satisfy such a condition. If the oscillators  503 ( 1 ) to  503 ( 3 ) are oscillated with the same phase, for example, as shown in  FIG. 27 , variable phase sections  102 ( 1 ) to  102 ( 3 ) for adjusting the phases need to be placed between the mixers  502 ( 1 ) to  502 ( 3 ) and amplifiers  104 ( 1 ) to  104 ( 3 ). A delay circuit may also be used as the variable phase section. 
   The resonators  1001 ( 1 ) to  1001 ( 3 ) extract signals according to the own resonance frequencies from the signals given from the amplifiers  104 ( 1 ) to  104 ( 3 ) and give them to the power combiner  106 . In this case, distortion resulting from non-linearity of active elements at the mixers  502 ( 1 ) to  502 ( 3 ) and amplifiers  104 ( 1 ) to  104 ( 3 ) are removed at the resonators  1001 ( 1 ) to  1001 ( 3 ) and the signals rid of distortion are given to the power combiner  106 . 
   The power combiner  106  combines the signals given from the resonators  1001 ( 1 ) to  1001 ( 3 ) to acquire a combined signal and outputs the combined signal acquired from an output terminal  92 . The combined signal outputted from the output terminal  92  is radiated into a space as an electric wave from an antenna (see  FIG. 24 ). The combined signal may be radiated from the antenna after passing through a filter. It is a great feature which is different from the configuration in  FIG. 24  that the distortion generated at the time of the frequency conversion by the mixers  502 ( 1 ) to  502 ( 3 ) is removed by the resonators  1001 ( 1 ) to  1001 ( 3 ) and not superimposed on the output signal. 
     FIG. 28  shows a 4-fold divider made up of two stages of 2-fold division Wilkinson type dividers using micro strip lines as a configuration example of the power divider  105 . 
   An input port  1  is provided at one end of a 50-ohm line  601  and one end of two 70.7-ohm 1/4-wavelength lines  602 ( 1 ),  602 ( 2 ) is connected to the other end of the 50-ohm line  601 . The other ends of the lines  602 ( 1 ),  602 ( 2 ) are connected each other by a 50-ohm resistor  603 , which constitutes a 2-fold divider. A 4-fold divider is realized by connecting these 2-fold dividers in two stages. One end of the 50-ohm lines  606 ( 1 ) to  606 ( 4 ) is connected to 1/4 -wavelength lines  605 ( 1 ) to  605 ( 4 ) in the second stage and output ports  2  to  5  are provided at the other ends of 50-ohm lines  606 ( 1 ) to  606 ( 4 ). Reference numerals  607 ,  608  denote 50-ohm resistors. Assuming that reference numerals  2  to  5  mean input ports and  1  mean an output port, the configuration shown in  FIG. 28  can be used as a combiner. 
     FIG. 29  shows a configuration example of a low-loss-oriented 4-fold divider at the sacrifice of an isolation characteristic. 
   An input port  11  is provided at one end of a 50-ohm line  601 , one end of each of four 100-ohm 1/4 -wavelength lines  604 ( 1 ) to  604 ( 4 ) is connected to the other end of the 50-ohm line  601 . 50-ohm lines  609 ( 1 ) to  609 ( 4 ) are connected to the other ends of the lines  604 ( 1 ) to  604 ( 4 ). Output ports  12  to  15  are provided on the output side of the 50-ohm lines  609 ( 1 ) to  609 ( 4 ). 
     FIG. 30  shows a configuration example where branching filters are used for the power divider  105  and resonance circuits  101 ( 1 ) to  101 ( i ) (assuming i=4) in  FIG. 1 . 
   A branching circuit  41  includes a resonance circuit  701  (corresponds to the resonator  101 ( 1 ) in  FIG. 1 ) which resonates at a resonance frequency f 1  at a termination section. The resonance circuit  701  resonates based on a signal inputted from ports  20  and the resonance signal is outputted from ports  21  through a coupling circuit  711  (corresponds to the coupling circuit which couples the resonator  101 ( 1 ) and variable phase section  102 ( 1 ) in  FIG. 1 ). A resonator  702  (corresponds to the resonator  102 ( 2 ) in  FIG. 1 ) of a resonance frequency f 2  is coupled with the branching circuit  41  at a position shifted from the position of the resonance circuit  701  of the resonance frequency f 1  by 1/4 wavelength (λ g2 /4) of the resonance frequency f 2 . The resonator  702  is coupled with a coupling circuit  712  (corresponds to the coupling circuit which couples the resonator  101 ( 2 ) with the variable phase section  102 ( 2 ) in  FIG. 1 ) and the resonance signal at the resonator  702  is outputted from ports  22  through the coupling circuit  712 . Likewise, a resonator  703  of a resonance frequency f 3  is coupled with the branching circuit  41  at a position shifted from the position of the resonance circuit  701  by 3/4 wavelength (3λ g3 /4) of the resonance frequency f 3 . The resonator  703  is coupled with a coupling circuit  713  (corresponds to the coupling circuit which couples the resonator  101 ( 3 ) with the variable phase section  102 ( 3 ) in  FIG. 1 ) and the resonance signal at the resonator  703  is outputted from ports  23  through the coupling circuit  713 . A resonator  704  of a resonance frequency f 4  is coupled with the branching circuit  41  at a position shifted from the position of the resonance circuit  701  by 5/4 wavelength (5λ g4 /4) of the resonance frequency f 4 . The resonator  704  is coupled with a coupling circuit  714  and the resonance signal at the resonator  704  is outputted from ports  24  through the coupling circuit  714 .