Abstract:
A power converter circuit is powered by a power source thereby generating a power line voltage. A rectifier is configured to rectify the power line voltage and generate a pulsating d.c. voltage with a frequency which is twice a frequency of the power line voltage. A gate drive circuit is designed to receive the pulsating d.c. voltage. The gate drive circuit drives a pair of switches which are controlled to invert the pulsating d.c. voltage, without use of level shifting. The gate drive circuit further includes a driving inductor, a timing inductor and a timing capacitor serially connected to each other and connected to the switches to control operation of the switches. The transformer, including a first transformer winding and a second transformer winding, and a capacitor are designed to receive an output signal from the switches. A lamp is connected to receive the driving signals from the transformer winding through a tap, where the tap attenuates the signal developed across the transformer winding.

Description:
FIELD OF THE INVENTION 
     The present invention relates to a ballast circuit for a low voltage halogen lamp, and more particularly, to such a ballast circuit employing a pair of complementary switches in a pulsating-d.c.-to-a.c. inverter. 
     BACKGROUND OF THE INVENTION 
     The typical method of supplying low voltage lamps, halogen in particular, from a high voltage supply (e.g., 120 volt, 60 Hz) is to use conventional half-bridge, high frequency inverters. These use a level shifting scheme to periodically switch on and off the high side switch. One design would be to employ a step-down transformer between the supply voltage and the low voltage lamp. Several disadvantages of this method are that transformers operating at 60 Hz are typically larger and heavier than would be needed at higher frequencies, step-down transformers are relatively expensive, and there is no inherent short circuit protection when employing a step-down transformer. 
     The present invention provides a ballast circuit for halogen, or other low voltage lamps, that overcomes the foregoing drawbacks. 
     BRIEF SUMMARY OF THE INVENTION 
     A ballast circuit is described for supplying high frequency a.c. current to a load circuit incorporating a low voltage halogen or other heated filament lamp and an autotransformer, or separate coupled transformers, and a capacitance. The ballast circuit includes an a.c.—d.c. converter circuit, comprising a full-wave rectifier and capacitance, and a pulsating-d.c.-to-a.c. inverter circuit comprising first and second switches serially connected between a bus conductor and a reference conductor. The switches are connected together at a common node through which the a.c. current flows, and at a control node. Also connected between the control node and the common node is a serially connected circuit including a capacitor and two inductors. The voltage between the control node and the common node is used to determine the conduction state of the associated switches, where the time constant of the serially connected circuit determines the frequency of operation. 
     Also included are first and second resistors connected serially between the bus conductor and the reference conductor and also connected to the control node. A third resistor is connected between the common node and one of the bus conductor and the reference conductor, to set the initial polarity of the control node with reference to the common node. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic representation of a preferred embodiment of the halogen power converter circuit according to the present invention; 
     FIGS. 2-7 illustrate alternate switches that can be used in the circuit of FIG. 1; 
     FIG. 8 is a waveform graph of a typical output voltage of the bridge rectifier (V BUS ); and 
     FIG. 9 is a graph of a typical voltage waveform at the halogen lamp (V LAMP ). 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1 depicts a power converter circuit  10  to drive a 12 volt halogen lamp  12 . Switches  14  and  16  are respectively controlled to convert pulsating d.c. current from a full-wave bridge rectifier  18 , connected to an a.c. source  20  (e.g. 120 volt, 60 Hz), to a.c. current received by a load circuit  22 , comprising an inductor/transformer winding  24  and a capacitor  26 . A pulsating d.c. bus voltage (V BUS ) exists between bus conductor  28  and reference conductor  30 . Load circuit  22  also includes halogen lamp  12 , which as shown is connected between node or tap  31  of transformer winding  24  and node  32 , which also connects transformer winding  24  to capacitor  36 . Capacitor  36  serves to maintain node  32  at about ½ bus voltage (V BUS ). A control circuit also called gate drive circuit  38  is provided, and in combination with switches ( 14 ,  16 ) form an inverter circuit configuration. 
     Capacitor  36  is connected across the bridge rectifier from bus conductor  28  to reference conductor  30 . Capacitor  36  is provided to prevent the pulsating d.c. voltage from going to zero between cycles. Capacitor  36  can, therefore, be much smaller than a typical filter capacitor (e.g., 0.1 μF). In this way, nearly unity power factor can be observed by the power line. By preventing the pulsating d.c. voltage from approaching zero, normal regenerative control circuit  38  operation is maintained, and the circuit does not need to be restarted. Control circuit  38  would operate satisfactorily if bus conductor  28  were supplied with pure d.c. current, however, this is not necessary for operation of the halogen power converter, and the expense of an electrolytic capacitor is avoided. 
     In power converter  10  of FIG. 1, switches  14  and  16  are complementary to each other in the sense, for instance, that switch  14  may be an n-channel enhancement mode device, and switch  16  may be a p-channel enhancement mode device as depicted in FIG.  1 . As shown in FIGS. 2 and 3 in one embodiment, each of switches  14  and  16  include an inherent, reverse-conducting diode  40  or  42 . When embodied as MOSFETs, each switch  14  and  16  has a respective gate, or control terminal. The voltage from the gate to source of switch  14  controls the conduction of that switch. Similarly, the voltage from the gate to source of switch  16  controls the conduction of that switch. As shown, the sources of switches  14  and  16  are connected together at a common node  44 . With the gates of switches  14  and  16  interconnected at common control node  44 , the single voltage between control node  46  and common node  44  controls the conduction state of both switches  14  and  16 . The drains of the switches are connected to bus conductor  28  and reference conductor  30 , respectively. 
     Switches  14  and  16  could alternatively be embodied as Insulated Gate Bipolar Transistor (IGBT) switches, such as the p-channel and n-channel devices respectively shown in FIGS. 4 and 5. However, each IGBT switch would then be accompanied by a reverse-conducting diode  48  or  50  as shown in FIGS. 4 and 5. An advantage of IGBTs over MOSFETs is that they typically have a higher voltage rating, enabling circuits with a wide range of d.c. input voltage values to utilize the same IGBTs. Further, switches  14  and  16  could be embodied as Bipolar Junction Transistor (BJT) switches, such as the NPN and PNP devices respectively shown in FIGS. 6 and 7. As with the IGBT switches, the BJT switches of FIGS. 6 and 7 are respectively accompanied by reverse-conducting diodes  52  and  54 . 
     Referring back to FIG. 1, gate drive circuit  38 , connected between control node  46  and common node  44 , controls the conduction states of switches  14  and  16 . Gate drive circuit  38  includes a driving inductor  58  that is mutually coupled to inductor transformer winding  24 , and is connected at one end to common node  44 . The end of transformer winding  24  connected to node  44  may be a tap from a transformer, formed by inductor windings  58  and  24 . Inductors  58  and  24  are poled in accordance with the solid dots shown adjacent the symbols for these inductors. Driving inductor  58  provides the driving energy for operation of gate drive circuit  38 . A second or timing inductor  60  and a timing capacitor  62  are serially connected to the remaining end of driving inductor  58 , between node  46  and inductor  58 . 
     A bi-directional voltage clamp  63 , comprising Zener diodes  64  and  66  in this embodiment, clamps positive and negative excursions of gate-to-source voltage to respective limits determined, e.g., by the voltage ratings of the back-to-back Zener diodes shown. The bi-directional voltage clamp is desirable where switches  14  and  16  comprise MOSFET or IGBT switches, however, where the switches comprise BJT switches, the bi-directional voltage clamp is not necessary and can be excluded. 
     A capacitor  68  is preferably provided between nodes  44  and  46  to predictably limit the rate of change of gate-to-source voltage between nodes  44  and  46 . This beneficially assures, for instance, a dead time interval in the switching modes of switches  14  and  16  wherein both switches are off between the times of either switch being turned on. 
     Resistors  70  and  72  are serially connected between bus conductor  28  and reference conductor  30 . A third resistor  74  is connected across switch  16  as shown. Resistors  70  and  72  are preferably of equal value if the duty cycles of switches  14  and  16  are equal. In such case, the average voltage during steady state at common node  44  is approximately ½ of bus voltage (V BUS )at bus conductor  28 , and setting the values of resistors  70  and  72  equal results in an average voltage at intermediate node  46  also of approximately ½ bus voltage (V BUS ). 
     Resistor  74  serves as a charging resistor, and is preferably connected between common node  44  and reference conductor  30  as shown in FIG.  1 . Resistor  74  may alternately be connected between common node  44  and bus conductor  28 . Resistor  74  aids in starting the regenerative control of first and second switches  14  and  16  during initial energization in the following manner. Upon energizing of source  20 , capacitor  62  becomes initially charged via resistors  70 ,  72  and  74 . At the instant that source  20  is energized, the voltage across capacitor  62  is zero, and, during the starting process, serially-connected inductors  58  and  60  act essentially as a short circuit due to the relatively long time constant for charging of capacitor  62 . If resistors  70 ,  72  and  74  are of equal value, for instance, the voltage on common node  44 , upon initial bus energizing, is approximately ⅓ of bus voltage (V BUS ). At this instant, the current flowing through resistor  70  is approximately equally divided between resistor  72  and the serially connected sequence of capacitor  62 , inductor  60 , inductor  58  and resistor  74 . In this manner, capacitor  62  becomes increasingly charged, from left to right as shown in FIG. 1, until it reaches the threshold voltage of the gate-to-source voltage of upper switch  14  (e.g., 2-3 volts). At this point, upper switch  14  switches into its conduction mode, which then results in current being supplied by that switch to load circuit  22 . In turn, the resulting current in the load circuit causes regenerative control of first and second switches  14  and  16  by means of inductive coupling between inductors  24  and  58 . Output capacitor  76  is connected in parallel to resistor  74 . 
     The frequency of operation in the present invention is not determined by a resonant load circuit as it is in prior art ballast circuits for gas discharge lamps. In the embodiment depicted in FIG. 1, the frequency of operation is determined by the time constant of serially connected inductors  58 ,  60  and capacitor  62 , respectively. The frequency of operation of control circuit  38  is, therefore, given by the following formula: 
     
       
         ƒ=1/2π C   t   L   g    (1),  
       
     
     where C t  is capacitor  62 , and L g  is inductor  60 . 
     The preferred frequency of operation for the halogen power converter is 30 kHz-90 kHz, however, this parameter is not critical. Note that the inverter frequency will change when the pulsating dc bus changes from the valleys of the power line to the crests. At the crests, the frequency is lowest, nearly 30 kHz, at the valleys, the inverter frequency is highest, nearly 100 kHz. This variation in the inverter frequency spreads the emissions out over a wide frequency range. Thus the inverter frequency is modulated by fluctuations in the bus voltage, which dramatically reduces the conducted EMI emissions on the power line. 
     The present invention satisfies at least five criteria for operating halogen lamps as enumerated below: 
     1. The output voltage is low (e.g., 12 volts). 
     2. The voltage crest factor is less than 2, which means that the peak output voltage does not exceed twice the rated lamp voltage (e.g., 24 volts for a 12 volt lamp). This ensures that no arcing occurs in the lamp. Also, this lower crest factor improves the efficiency of the inverter circuit. 
     3. The power converter does not reduce the life of the lamp. 
     4. The power converter is short circuit proof. 
     5. The power converter is inexpensive. 
     With regard to the first criterion, FIG. 1 shows the lamp  12  connected to the low voltage winding  24  of the autotransformer formed by winding  24  and winding  58 . 
     An explanation as to the satisfaction of the second criterion is provided below: 
     As discussed earlier, bus conductor  28  has a pulsating d.c. voltage impressed upon it as shown in FIG.  8 . The waveform in FIG. 8 is nonzero in the valleys due to the use of capacitor  36  as previously noted. For purposes of explanation it will be assumed that a.c. source  20  is a typical 120 volt, 60 Hz sinusoidal waveform. Therefore, the peak voltage is 120{square root over (2)}≈169.7 volts. The regenerative cycling of control circuit  38  alternately connects node  44  to bus conductor  28  and reference conductor  30  at approximately 30 kHz at the peak of the line and approximately 90 kHz at the nonzero valleys. Therefore, the waveform appearing on the upper connection of autotransformer ( 24 ,  58 ) is approximately a 30-90 kHz square wave with a 120 Hz sinusoidal envelope impressed upon it as illustrated graphically in FIG.  9 . Please note that, for readability, FIG. 9 shows a 3 kHz waveform inside a 120 Hz envelope. The ratio of the series winding of autotransformer ( 24 ,  58 ) to the common winding to which lamp  12  is connected is provided via tap  31 , and is selected such that the lamp voltage is, for example, 12 volts RMS for a 12 volt lamp. Since the envelope is a sinusoidal waveform, the peak voltage seen by lamp  12  will be 12{square root over (2)}≈17 volts. The crest factor is, therefore, {fraction (17/10)}=1.7 which is less than 2 by a comfortable margin. 
     The third criterion is satisfied simply by ensuring that the RMS lamp voltage appearing on the common winding of autotransformer ( 24 ,  58 ) matches the rated lamp voltage (e.g., 12 volts), and also ensuring that the crest factor is less than 2 as explained previously. This prevents damaging arcing from occurring within the lamp. 
     The fourth criterion is satisfied because of the increased leakage inductance at autotransformer ( 24 ,  58 ). In typical prior-art power converters, line voltage is simply reduced to from 120 volts to 12 volts with a step-down transformer. The leakage inductance of a step-down transformer is insufficient to prevent damagingly high currents from flowing in the event of a short circuit across the transformer&#39;s secondary winding. In the halogen power converter of the present invention, there is adequate leakage inductance at the preferred 30 kHz to limit current to safe values. 
     The fifth criterion is satisfied for the following reasons. Because the transformers and inductors are operating at 30 kHz, as compared to 60 or 120 Hz for example, the transformers and inductors can be of a much smaller size as is well known in the art. Further, no electrolytic filter capacitor is required for satisfactory operation of control circuit  38 . Capacitor  36  is a small capacitor (e.g., 0.1 μF), and only needs to be large enough to sustain regenerative operation of control circuit  38  in the valleys of the pulsating d.c. current supplied on bus conductor  28 . Current flow through lamp  12  at that time is very small, and, consequently, capacitor  36  can be very small. 
     Thus, the present invention describes a power converter  10  for driving a 12 volt lamp, where power line voltage  20  is rectified and fed to the bus conductor  28  of the inverter ( 38 ,  14 ,  16 ), so no smoothing capacitor is needed. By this design, a pulsating d.c. waveform is applied to the bus conductor  28  whose fundamental frequency is twice the power line frequency. The pulsating d.c. is inverted by the complementary pair of switches ( 14 ,  16 ) that require no level shifting. They are driven from node  44 . Tap  31  attenuates the inverter voltage developed across autotransformer ( 24 ,  58 ) to about 12 volts rms. This low cost method powers the 12 volt lamp at high frequency, thus reducing the size of the inductive components. It is to be understood that the disclosed power converter may be used as an integral part of a lamp system or as an external power converter. 
     Exemplary component values for the circuit of FIG. 1 are as follows for a lamp rated at 12 volts, with a supply voltage of 120 volts a.c.: 
     
       
         
               
               
               
               
             
           
               
                   
                   
               
             
             
               
                   
                 Load inductor (24) 
                 10 
                 millihenries 
               
               
                   
                 Turns ratio between (24) and (58) 
                 3 
               
               
                   
                 Load capacitor (26) 
                 0.22 
                 microfarads 
               
               
                   
                 Capacitor (36) 
                 0.22 
                 microfarads 
               
               
                   
                 Driving inductor (58) 
                 75 
                 microhenries 
               
               
                   
                 Timing inductor (60) 
                 1 
                 millihenries 
               
               
                   
                 Timing capacitor (62) 
                 22 
                 nanofarads 
               
               
                   
                 Zener diodes (64, 66) each 
                 10 
                 volts 
               
               
                   
                 Capacitor (68) 
                 2.2 
                 nanofarads 
               
               
                   
                 Resistors (70, 72, 74) each 
                 270 
                 k ohms 
               
               
                   
                 Capacitor (76) 
                 470 
                 picofarads 
               
               
                   
                   
               
             
          
         
       
     
     While the invention has been described in conjunction with specific embodiments thereof, it is evident that many alternatives, modifications, and variations will be apparent to those skilled in the art in light of the foregoing description. Accordingly, it is intended to embrace all such alternatives, modifications, and variations as fall within the spirit and broad scope of the appending claims.