Abstract:
A direct conversion receiver uses an algorithm implemented by a DSP to cancel residual DC offsets during demodulation of a GMSK modulated signal. The algorithm exploits the characteristics of GMSK modulation by determining the. modulation extremes within sampled I/Q signals and calculates the DC offset as the mean of the extremes. This offset is used to weight a declining exponential function which is subtracted from the original signal samples to achieve compensation.

Description:
FIELD OF THE INVENTION 
   This invention relates to the field of direct conversion radio receivers, particularly but not exclusively to a method of DC compensation in a direct conversion receiver. 
   BACKGROUND 
   Direct conversion radio receivers, also referred to as zero-IF receivers, inherently suffer from the problem of DC offsets. Sources of DC offset are present in all stages of a direct conversion receiver due to component imbalances and tolerances. Further sources of DC offset include leakage of the local oscillator signal used in the receiver, for example, conduction and/or radiation of the local oscillator signal to the receiver antenna and leakage of the received signal into the local oscillator port of the RF mixers used in the receiver. 
   In a conventional direct conversion radio receiver operating, for example, in accordance with the GSM standard, which uses GMSK (Gaussian Minimum Shift Keying) modulation, the received signal is coupled to RF mixers which provide a baseband in-phase signal component I and a baseband quadrature signal component Q. 
   A hardware solution to the DC offset problem is implemented in direct conversion receivers such as that used in the Nokia 6210™ mobile telephone. The fact that the receiver is intended for use with GSM time division multiple access (TDMA) signals has been exploited to incorporate time periods in which the DC component of the I/Q signals can be clamped to zero by a DC cancellation (DCN) circuit. The timing of the DCN periods is controlled to occur in advance of the reception of any received signal burst. The DCN circuits are designed as high pass filters, in which capacitors can be rapidly charged/discharged during the DCN period by electronic switching circuits, to obtain a subtraction of the DC offset in each I or Q channel. The weakness of this solution is that the precision of the DC compensation is dependent on the RF signal received during the DCN period. Because the DCN period lies in the time slot preceding the time slot allocated for the mobile station&#39;s own received signal, the signal in the DCN period can range from OdBm to, for example, more than 30 dB higher than the level in the receive time slot allocated to the mobile station. When the received level in the DCN period is, for example, 30 dB higher than the mobile station&#39;s own received signal, the GMSK modulation cannot be averaged sufficiently by the DCN capacitors, resulting in an excessive residual DC content in the I/Q samples of the mobile station&#39;s own signal. This in turn results in a poor bit error rate (BER) for the received burst. 
   The above described problem also prevents direct conversion receivers using the DCN hardware solution from complying with certain tests used in GSM Type Approval, where signals which differ by 20 dB between adjacent time slots are applied to a mobile station. 
   The present invention aims to address the above problems. 
   SUMMARY OF THE INVENTION 
   According to the present invention, there is provided a method of DC compensation for a direct conversion radio receiver, comprising the steps of determining the modulation extremes of a received modulated signal, determining a DC offset for the signal from the modulation extremes; and processing the signal to compensate for the offset. 
   The DC offset can be determined as substantially the mean of the signal amplitude at the modulation extremes. The modulation extremes can be determined from an inverse filtered signal, where the inverse filtering operation is applied to compensate for the response characteristic of the baseband circuit. 
   According to the present invention, there is further provided a direct conversion receiver comprising means for determining the modulation extremes of a received modulated signal, means for determining a DC offset for the signal from the modulation extremes; and means for processing the signal to compensate for the offset. 
   According to the present invention, there is also provided a program to be executed by a digital signal processor in a direct conversion receiver, the receiver comprising a mixer circuit for providing quadrature related signals from a received modulated signal, a dc cancellation circuit for cancelling the dc component in the quadrature related signals and a digital signal processor for removing a residual dc component from the signals, said program being configured to cause the digital signal processor to determine the modulation extremes of the signals, to calculate a dc offset for the signals from the modulation extremes and to process the signals to compensate for the dc offset. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Embodiments of the invention will now be described, by way of example, with reference to the accompanying drawings, in which: 
       FIG. 1  is a schematic block diagram illustrating a mobile telephone handset and its associated network; 
       FIG. 2  is a schematic block diagram showing the mobile handset of  FIG. 1  in more detail; 
       FIG. 3  is a block diagram of a direct conversion radio receiver which forms the RF stage shown in  FIG. 2 ; 
       FIG. 4  is a flow diagram illustrating the DC cancellation algorithm according to the invention; 
       FIG. 5  shows a distorted I signal; 
       FIG. 6  shows a filtered I signal; 
       FIG. 7  shows the I signal after subtraction of a compensating exponential function from the sampled signal; 
       FIG. 8   a  is a scattering diagram showing I/Q samples before applying compensation in accordance with the invention; and 
       FIG. 8   b  is a scattering diagram showing the I/Q samples of  FIG. 8   a  after applying compensation in accordance with the invention. 
   

   DETAILED DESCRIPTION 
   In  FIG. 1 , a mobile station in the form of a battery driven telephone cellular handset MS 1 , is shown schematically in radio communication with a public land mobile network PLMN  1 . 
   The mobile handset MS 1  includes a microphone  2 , keypad  3 , further keys comprising soft keys  4   a ,  4   b  and a navigation key  4   c , a liquid crystal display  5 , earpiece  6  and internal antenna  7 . The handset  1  is WAP-enabled. The Nokia 6210™ is an example of a WAP-enabled mobile handset. 
   The circuitry of the handset  1  is shown in more detail in  FIG. 2 . Signal processing is carried out under the control of a digital micro-controller  8  which has associated RAM  9  and flash memory  10 . Electrical analogue audio signals are produced by microphone  2  and amplified by pre-amplifier  11 . Similarly, analogue audio signals are fed to the earpiece  6  through an amplifier  12 . The micro-controller receives instruction signals from the keypad  3 , soft keys  4   a ,  4   b  and navigation key  4   c  and controls operation of the LCD display  5 . The soft-keys  4   a ,  4   b  comprise user-programmable keys, while the navigation key  4   c  comprises, for example, a roller device to perform a scrolling function for the display. 
   Information concerning the identity of the user is held on a smart card  13  in the form of a GSM SIM card which contains the usual GSM international mobile subscriber identity and encryption K, that is used for encoding the radio transmission in a manner well known per se. The SIM card  13  is removably received in a SIM card holder  14 . Radio signals are transmitted and received by means of the antenna  7  connected through an RF stage  15  to a codec  16  configured to process signals under the control of the micro-controller  8 . 
   Thus, in use, for speech, the codec  16  receives analogue signals from the microphone amplifier  11 , digitises them into a form suitable for transmission and feeds them to the RF stage  15  for transmission through antenna element  7  to the PLMN  1  shown in  FIG. 1 . Similarly, signals received from the PLMN  1  are fed through the antenna element to be demodulated by the RF stage  15  and fed to codec  16  so as to produce analogue signals fed to amplifier  12  and earpiece  6 . 
   Referring to  FIG. 3 , the RF stage  15  includes a direct conversion receiver, which comprises a low noise amplifier (LNA)  20 , which receives input signals from the antenna  7  and the output of which is connected to a mixer circuit  21  which comprises first and second mixers  22 ,  23 , a local oscillator  24  and a 90° phase shifter  25 . The output of the LNA  20  is connected to respective first inputs of the first and second mixers  22 ,  23 . The output signal from the local oscillator  24  is connected to a second input of the first mixer  22  and, via the 90° phase shifter  25 , to a second input of the second mixer  23 . The respective outputs  26 ,  27  of the mixer circuit  21  are connected to a baseband circuit  28 . The baseband circuit  28  comprises respective baseband amplifiers  29 ,  30 , low pass filters  31 ,  32 , DC cancellation circuits  33 ,  34  and analog-to-digital converters  35 ,  36 . The respective outputs  37 ,  38  of the baseband circuit are fed to a digital signal processor  39 . 
   The operation of the direct conversion receiver will now be described in detail. As mentioned above, GSM uses a modulation scheme known as Gaussian Minimum Shift Keying (GMSIK). Reference is directed to Mouly &amp; Pautet; “The GSM System for Mobile Communications”, pages 249 to 259, for a detailed discussion of the GMSK modulation system for GSM. GMSK modulated signals are received at the antenna  7 , amplified by the low noise amplifier  20  and fed to the mixer circuit  21 . The mixer circuit  21  uses the well-known technique of multiplying the modulated carrier with the local oscillator signal provided by the local oscillator  24  which has the same frequency as the carrier wave. The resulting outputs  26 ,  27  comprise a first signal  26  referred to as the in-phase channel I and a second signal  27  referred to as the quadrature channel Q. Taking the example of the I signal, this is amplified by a baseband amplifier  29  and filtered by a low pass filter  31 . DC cancellation is then applied by a DC cancellation circuit (DCN)  33 , and the resulting signal is digitised by the analog-to-digital converter  35  and fed to the DSP  39 . The Q signal is processed in an exactly analogous way which is therefore not described separately. As mentioned previously, the DCN circuits  33 ,  34  are effectively high-pass filters, in which capacitors can be rapidly charged/discharged by electronic switching circuits during selected DCN periods, to obtain a subtraction of the DC offset in each of the I and Q channels. For example, during a DCN period, the received signal is used to rapidly charge a capacitor in the DCN circuit to a value which represents the DC content of the signal, so that the output of the DCN circuit during the following burst period, when the DCN function is deactivated, is compensated by the offset measured during the DCN period. 
   The I and Q samples received by the DSP  39  are still distorted by residual DC content as a result of insufficient DC cancellation in the DC cancellation circuits  33 ,  34 . Additional DC compensation is therefore provided in the DSP software, which implements a DC cancellation algorithm, shown schematically in  FIG. 4 . 
   A characteristic of GMSK modulation is that for a limited number N of I/Q samples, there is a sufficiently high probability that the I/Q vector has been positioned in all of the possible constellation points, or in other words that both the I and Q signals will have been in their modulation extremes. A modulation extreme results from a large number of successive repetitions of the same symbol in the original NRZ (Non-Return-to-Zero) sequence to be transmitted. By finding the modulation extremes over N samples for each of the I and Q channels, the DC content can be calculated as the mean of the two extremes. 
   In the example described above, the receiver hardware is found to exhibit a high pass filter characteristic. As a result, the DC component in this example is a declining function; the DC content being largest at the start of the burst and zero at the end of the burst. 
     FIG. 5  shows a distorted I signal from which the modulation extremes need to be determined. Referring to  FIGS. 4 and 5 , the signal is first filtered to straighten it, by applying an inverse filter to counteract the effect of the hardware high-pass filter characteristic (step s 1 ). 
   Assuming that the high pass characteristics of the hardware can be modelled as a first order high-pass filter of the form: 
             H   ⁡     (   s   )       =     s     s   +     1   /   RC               
where RC denotes the effective RC product of the receiver circuit, then the inverse filter is implemented in the digital domain as:
 
             S   ⁡     (   z   )       =       1   -     τ   ⁢           ⁢     z     -   1             1   -     z     -   1                 
where τ is a direct matched z—transform of the zero which cancels the single pole in H(s), and
 τ= e   −1/RC*fs    
where fs is the sample frequency.
 
   The inverse filter is implemented in the DSP  39  as:
 
 y=x−τxz   −1   +yz   −1  
 
   The constant τ is tuned to a value representative of the components used in the direct conversion receiver hardware. With normal component tolerances, the value need only be determined once for each particular type of mobile station. For example, for the Nokia 6210™, the following values are used:
 
 RC= 1.349×10 −4 , τ=0.973
 
     FIG. 6  shows the result of applying the inverse filter to the signal shown in  FIG. 5 , at a sample rate which equals the symbol rate. In this example, only the first 64 samples in the burst are inverse filtered, as indicated by the window  40 . 64 samples have been found to be a sufficiently high number to obtain the required probability of finding the modulation extremes. Although it is possible to scan a longer part of the burst to increase the chances of finding the true modulation extremes, such scanning requires a very accurate match of the inverse filter. If the match is not very accurate, a drift in the DC level can occur which increases for samples taken progressively further from the start of the burst. 
   The modulation extremes are then determined (step s 2 ), being the signal amplitude values I_max and I_min within the 64 sample window  40  shown in  FIG. 6 . The determined values are used to calculate the DC offset for the I channel (step s 3 ): 
   
     
       
         
           I_offset 
           = 
           
             
               I_max 
               + 
               I_min 
             
             2 
           
         
       
     
   
   A similar procedure is carried out for the Q channel, resulting in a value of the offset for the Q channel (step s 3 ): 
   
     
       
         
           Q_offset 
           = 
           
             
               Q_max 
               + 
               Q_min 
             
             2 
           
         
       
     
   
   As mentioned above, the high pass characteristics of the receiver hardware lead to the DC component in this example being a declining exponential function, so that compensation is performed by subtracting a weighted declining exponential function from the I/Q samples in the DSP, the algorithm providing an accurate determination of the initial DC content in the samples in the received burst and therefore the weighting for the exponential function. 
   The value of τ determined above is also used for the calculation of the declining exponential function used for compensation of the I/Q samples (step s 4 ): 
   Exponential function:
 
 E   n   =E   n −1·τ
 
where E 0 =1
 
   Values to be subtracted from I samples:
 
 IO   n   =I _offset· E   n  
 
   The compensated I signals are determined by subtracting the weighted declining exponential function values from the original I samples (step s 5 ):
 
 I   comp   =I   n   −IO   n  
 
     FIG. 7  shows the compensated I signal I comp  after subtraction of IO n  from the samples. 
   The Q signal is treated in an entirely analogous way to provide a compensated Q signal Q comp . 
     FIG. 8   a  is a scattering diagram of the I/Q samples without applying compensation in accordance with the invention.  FIG. 8   b  is a scattering diagram of the same samples after processing by the algorithm, illustrating the significant decrease in sample scattering observed in accordance with the invention. 
   In principle, it is also possible to subtract the determined DC offset values from the inverse filtered signal to obtain the final compensated I/Q signals, which can also enable the bandwidth reduction introduced by the hardware high-pass filter to be eliminated. 
   While the above example has been described in relation to a receiver having a first order high-pass characteristic, the receiver can have a baseband characteristic of a higher order. For such a receiver, a corresponding higher order inverse filter is applied in the DSP in order to straighten the signal to enable the modulation extremes to be determined. 
   Furthermore, for receivers which do not exhibit a high pass characteristic, there is a constant residual DC content in the signal received during the burst period. In this case, the inverse filtering operation is not required and compensation is achieved by subtracting a fixed DC level, rather than a declining exponential DC-level, from the received samples. 
   It will be appreciated that, once compensated I/Q values are available, subsequent processing is applied to correctly demodulate the GMSK signals and recover the original transmitted data, in a manner which is well known per se. 
   While the invention has been described in relation to GMSK modulation, the skilled person would appreciate that the principles could be applied to various other types of modulation, in particular constant amplitude modulation schemes, including for example the proposed 8 PSK modulation scheme in the GSM EDGE evolution.