Abstract:
A frequency upconverter using mixers operating on one or more signals and inverted versions thereof and a subtractor, such as a balun, for subtractively combining the mixer outputs to produce an upconverted signal.

Description:
FIELD OF THE INVENTION 
     The invention relates to frequency conversion processes, such as are used in radio communications systems. 
     DESCRIPTION OF RELATED ART 
     Information is exchanged in a radio communication system via a modulated radio frequency (RF) carrier wave. Typically, the modulating signal varies either the amplitude or the phase of the carrier wave (called AM and PM, respectively) although in modern communication systems both amplitude and phase are often varied simultaneously as in Quadrature Amplitude Modulation (QAM). 
     A simple (and therefore cost-effective) way of generating a modulated carrier is by a process known as direct upconversion. Here, a baseband version of the modulated signal is translated from baseband to RF in a single step. In a modern radio communication system, the baseband modulated signal is normally generated digitally and converted into the analogue domain with the use of a digital-to-analogue converter (DAC). If the modulated signal has varying phase and amplitude, then the baseband signal is defined in terms of two independent signals that together define a vector that determines the signal&#39;s amplitude and phase. The two independent signals are often specified in the cartesian coordinate system and called I and Q. These define the amplitude of two carrier waves that have a 90° phase difference between them. It is usual to provide a pair of DACs, one for producing each of the I and Q baseband signals. Often, the I and Q signals are provided in differential form. 
     The device normally used for direct upconversion is called a quadrature modulator and is shown in  FIG. 1 . The quadrature modulator of  FIG. 1  comprises two mixers  10 ,  12 , which operate with quadrature local oscillator (LO) signals  14 ,  16  having a 90° phase difference between them. The I and Q baseband signals  13 ,  15  are produced as differential signals by respective DACs (not shown). The I baseband signal  13  is applied to one mixer  10  and the Q baseband signal  15  is applied to the other mixer  12  such that the RF mixer outputs  18 ,  20  are amplitude-modulated versions of the LO signal. The RF outputs  18 ,  20  of the two mixers are then combined at summer  22  to produce the modulated RF carrier  24 . 
     The main difficulty with direct upconversion is the generation of spurious signals that are often so close (spectrally) to the wanted signal that they cannot be removed by simple filtering at the modulator output.  FIG. 2  shows typical spurious signals for the modulator of  FIG. 1 . The wanted signal is indicated at  25 . The spurious signals of greatest concern are, first, a signal  26  at the LO frequency which is caused by leakage of the LO signal across the mixers  10 ,  12  to their RF outputs  18 ,  20  and, second, an ‘image’  28 ,  30  caused by differential gain and phase errors in the I and Q paths. 
     It can be shown that these particular spurious signals can be removed at the modulator output  24  by modifying the baseband signal applied to the modulator input. This is readily done in the digital domain prior to analogue conversion. The LO leakage signal  26  is removed by adding a dc signal to each I and Q baseband signal such that an additional signal component at the LO frequency appears at the modulator output which has the same amplitude but opposite phase to the LO leakage signal  26 . Hence, the LO leakage signal is removed by cancellation. The ‘image’  28 ,  30  is removed by the addition of digital equalisation filters into each I and Q path in advance of the DACs. These filters compensate for the gain and phase ripple and offsets in each path and thereby remove the spurious ‘image’ signal. 
     A significant problem with digital compensation of LO leakage is the loss of dynamic range (signal-to-noise ratio) from the DAC if the LO leakage signal  26  is too large. This is because the DAC must produce a large dc compensating signal in addition to the wanted signal. This moves the origin of the baseband signal away from zero and reduces the dynamic range available for the wanted baseband signal. This is illustrated in  FIG. 3 , which will now be discussed. 
     The vectors X and Y indicate, respectively, the maximum values that can be produced by the DACs that produce the I and Q baseband signals. Therefore, the DAC outputs can specify any point in rectangle  32 , which ranges from −|X| to |X| and from −|Y| to |Y|. Commonly, the modulation scheme employed will dictate that the sum of the squares of the moduli of the I and Q baseband signals shall not exceed a maximum value. This has the effect of confining the outputs of the DACs to specifying a point within a circular area within the aforementioned rectangle. 
     Ordinarily, this circle would be centered upon the origin of the vectors X and Y, which is also the center point of the rectangle  32 . In such circumstances, circle  34  represents the largest circle that can be fitted within rectangle  32 , and hence the radius of circle  34  indicates the maximum signal level and hence dynamic range that can then be given to the baseband signal. When dc offsets are applied to the I and Q baseband signals to cancel the LO leakage signal  26 , the center point of the circle confining the DAC outputs is displaced. In  FIG. 3 , this displacement is indicated by vector A LO , which is the vector sum of the dc offsets. With this new center point, the maximum circle that can be placed within rectangle  32  is circle  36 , which is clearly of lesser radius than circle  34  indicating a reduction in the maximum dynamic range of the baseband signal. 
     Another problem with direct upconversion is distortion of the baseband signal, which results in spurious signals at the modulator output that are often too close (spectrally) to the wanted signal  25  to be easily removed by filtering after the modulator.  FIG. 4  illustrates the appearance of three spurious signals  40 ,  42 , and  44  in the vicinity of the wanted signal  25  due to second-order baseband distortion. This distortion can be generated anywhere in the baseband circuit and is often produced in the DACs, the mixers, and baseband buffers (if any). 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       By way of example only, the invention will now be described with reference to the accompanying drawings, in which: 
         FIG. 1  illustrates a prior-art, quadratic modulator architecture with differential baseband inputs; 
         FIG. 2  illustrates spurious signals appearing at the output of the modulator of  FIG. 1  due to LO leakage and differential gain and phase errors between the I and Q channels; 
         FIG. 3  illustrates permissible ranges of DAC outputs for the modulator of  FIG. 1 ; 
         FIG. 4  illustrates spurious signals due to second-order baseband distortion at the output of the modulator of  FIG. 1 ; 
         FIG. 5  illustrates a push-pull upconverter; and 
         FIG. 6  illustrates the relative direction of signal vectors in the I channel of the upconverter of  FIG. 5 . 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     According to one aspect, the invention provides apparatus for frequency upconversion of a signal comprising two components, the apparatus comprising two pairs of mixers, each pair adapted to act on a respective one of the components and, in each pair, one mixer adapted to upconvert the component assigned to that pair and the other mixer adapted to upconvert an inverted version of that component; a pair of subtractors, each subtractor adapted to combine by subtraction the outputs of a respective one of the pairs of mixers; and a combiner adapted to combine the outputs of the subtractors to produce an upconverted version of the signal. 
     The invention also consists in a method of frequency upconversion of an input signal comprising two components, the method comprising a first upconverting step comprising mixing one of the components with a conversion signal to produce a first upconverted signal and mixing an inverted version of that component with the conversion signal to produce a second upconverted signal; a second upconverting step comprising mixing the other of the components with a conversion signal to produce a third upconverted signal and mixing an inverted version of that component with the conversion signal to produce a fourth upconverted signal; a first subtracting step comprising combining by subtraction the first and second upconverted signals to produce a first difference signal; a second subtracting step comprising combining by subtraction the third and fourth upconverted signals to produce a second difference signal; and a combining step comprising combining the first and second difference signals to produce an upconverted version of the input signal. 
     According to another aspect, the invention provides apparatus for frequency upconversion of a signal, the apparatus comprising a pair of mixers, one mixer adapted to upconvert the signal and the other mixer adapted to upconvert an inverted version of the signal; and a subtractor adapted to combine by subtraction the outputs of the mixers to produce an upconverted version of the signal. 
     The invention also consists in a method of frequency upconversion of an input signal, the method comprising an upconverting step comprising mixing the input signal with a conversion signal to produce a first upconverted signal and mixing an inverted version of the input signal with the conversion signal to produce a second upconverted signal; and a subtracting step comprising combining by subtraction the first and second upconverted signals to produce an upconverted version of the input signal. 
     The invention also extends to apparatus for frequency upconversion of an input signal, the apparatus comprising an oscillator adapted to provide a conversion signal, a pair of mixers, one mixer adapted to upconvert the input signal with the conversion signal and the other mixer adapted to upconvert an inverted version of the input signal with the conversion signal; and a combiner adapted to combine the outputs of the mixers to produce an upconverted version of the signal in which leakage of the conversion signal through the mixers is substantially suppressed. 
     The invention also relates to apparatus for frequency upconversion of a signal, the apparatus comprising a pair of mixers, one mixer adapted to upconvert the signal and the other mixer adapted to upconvert an inverted version of the signal; and a combiner adapted to combine the outputs of the mixers to produce an upconverted version of the signal in which even-order distortion created by the mixers is substantially suppressed. 
     The invention also provides apparatus for frequency upconversion of a signal, the apparatus comprising two processing paths, each path comprising an amplifier and a mixer, wherein the amplifier and mixer in one path are respectively adapted to amplify and upconvert the signal, and the amplifier and mixer in the other path are respectively adapted to amplify and upconvert an inverted version of the signal; and a combiner adapted to combine the outputs of the paths to produce an upconverted version of the signal in which even-order distortion created by the amplifiers is substantially suppressed. 
     In the various aspects mentioned above, the invention utilises subtraction in the production of the upconverted signal. In certain embodiments, this subtraction is performed by a balun that combines two signals whilst, at the same time, rotating one of them by 180° in phase. 
     Moreover, the various aspects of the invention discussed above involve signal inversion. Where a signal to be inverted is a differential signal then, in certain embodiments, the inversion is achieved by swapping the components of the differential signal. 
     The upconverter of  FIG. 5  is called a push-pull upconverter because of its ability to cancel out certain errors that are introduced by its mixers. The push-pull upconverter of  FIG. 5  has certain similarities with the modulator of  FIG. 1 . The main differences are that the push-pull upconverter uses a pair of mixers in place of each of the mixers  10 ,  12  of  FIG. 1  and that the RF outputs of the mixers are combined in baluns. 
     Unlike  FIG. 1 ,  FIG. 5  illustrates the DACs that produce the I and Q baseband signals. DAC  46  produces a differential I baseband signal  48  comprising an I+ signal and an I− signal. DAC  50  produces a differential Q baseband signal  52  comprising Q+ and Q− signals. The differential I and Q baseband signals  48 ,  52  are supplied to respective reconstruction filters  54  and  56 . The reconstruction filters smooth the relatively rough waveforms of the differential I and Q baseband signals as produced by the DACs  46  and  50 . 
     After passage through the reconstruction filters, each of the I+, I−, Q+, and Q− signals is supplied to a respective resistive splitter  58 ,  60 ,  62 , and  64 . The splitters pass their outputs to four mixers  66 ,  68 ,  70 , and  72 . Each of the mixers has, as shown, an IF+ and an IF− port, each for receiving a part of a differential signal. Mixer  66  receives the signal I+ at its IF+ port from splitter  58  and receives the signal I− at its IF− port from splitter  60 . Mixer  66  therefore receives the differential I baseband signal. Mixer  68  receives the signals I− and I+ at its IF+ and IF− ports, respectively, from splitters  60  and  58 , respectively. Mixer  68  therefore receives a version of the differential I baseband signal that has been phase-shifted by 180° relative to the version that is supplied to mixer  66 . Similarly, mixer  70  receives the signals Q+ and Q− at its IF+ and IF− ports, respectively, from splitters  62  and  64 , respectively, and mixer  72  receives the signals Q− and Q+ at its IF+ and IF− ports, respectively, from splitters  64  and  62 , respectively. Therefore, mixer  72  receives a version of the differential Q baseband signal that is in antiphase relationship with the version of the differential Q baseband signal that is supplied to mixer  70 . 
     A local oscillator (LO) signal  74  is supplied to a quadrature splitter  76 , which emits two versions  78 ,  80  of the LO signal  74 . Signal  80  is a version of the LO signal  74  that has been retarded in phase by 90°, and signal  78  is a version of the LO signal  74  with no phase offset. Signal  78  is applied to mixers  66  and  68  via a further splitter  82 , and signal  80  is applied to each of mixers  70  and  72  via a further splitter  84 . Hence, mixers  66  and  68  mix oscillator signal  72  with antiphase versions of the differential I baseband signal, and mixers  70  and  72  mix oscillator signal  80  with antiphase versions of the differential Q baseband signal. 
     RF signals  86  and  88  produced by mixers  66  and  68  are supplied to a balun  90 . Balun  90  combines the wanted, modulated signals in phase and combines the LO leakage signals in antiphase such that they cancel. Furthermore, any even-order baseband distortion generated in mixers  66  and  68  or in any baseband buffer amplifiers placed just before the mixers also combines in antiphase and cancel. Balun  90  can be said to subtract one of signals  86  and  88  from the other. The action of balun  90  can be explained mathematically, as follows. 
     If we consider the pair of mixers  66  and  68  as having baseband inputs defined as I 1  and I 2 , we can write the output from balun transformer  90  as:
 
 V   ol ( t )= I   1 ( t )cos(ω LO   t )− I   2 ( t )cos(ω LO   t )
 
If we now define the LO leakage from each mixer as l cos(ω LO t+α) and add a 2 nd  order (AM-AM) distortion product to the baseband inputs and let I 2 (t)=−I 1 (t), we obtain:
 
 V   ol ( t )=( I   1 ( t )+ cI   1 ( t ) 2 )cos(ω LO   t )+ l  cos(ω LO   t +α)−(− I   1 ( t )+ cI   1 ( t ) 2 )cos(ω LO   t )− l  cos(ω LO   t +α)
 
The LO leakage signal and the even order distortion products cancel and we obtain:
 
 V   ol ( t )=2 I   1 ( t )cos(ω LO   t )
 
as required.
 
     The above mathematical description is supplemented by the diagrammatic representation provided in  FIG. 6 , which illustrates the relative direction of the signal vectors at various points in the path leading to balun  90 .  FIG. 6  also illustrates, at  85  and  87 , how buffer amplifiers can be included upstream from the mixers. 
     RF outputs  92  and  94  of mixers  70  and  72  are supplied to another balun  96 . Balun  96  operates in the same way as balun  90 . That is to say, balun  96  combines the wanted modulated signals in phase, the LO leakage signals in antiphase, and any even-order baseband distortion generated by mixers  70  and  72 , or in any baseband buffer amplifiers placed just before them, in antiphase. 
     The outputs of baluns  90  and  96  are combined by a summer  98  which produces the quadrature modulator output V m . 
     Relative to RF output  86  or  88  of mixer  66  or  68 , the wanted signal power at the output of balun  90  is increased by 3 dB, the LO leakage signal is reduced, and spurious signals due to even-order baseband distortion generated after the resistive splitters is reduced. 
     In practice, the baluns will not perform perfect cancellation of the targeted spurious signals. This is because a balun will not phase-shift one of its inputs relative to the other by precisely 180°. Another reason is that the mixers that supply RF signals to a given balun will not be precisely the same, leading to a difference in the LO leakage signals that they produce. Hence, even in the circuit of  FIG. 5 , there often remains a need to introduce dc correction signals to the I and Q baseband signals in the digital domain for the correction of the LO leakage signal. However, the size of the vector A LO  will be much smaller than before, leading to a substantial improvement in available dynamic range. 
     For a given modulator output power, each mixer in  FIG. 5  operates with a 3 dB-lower output power than the mixers in  FIG. 1 . Thus, spurious signals generated by non-linearities in the mixers, and in any buffer amplifiers upstream of the mixers, will be reduced compared to what would be generated in  FIG. 1  with mixers of the same type. This reduction in the level of spurious signals is in addition to the cancellation in the baluns of spurious signals due to even-order baseband distortion. 
     It should also be noted that the highly balanced nature of the circuit of  FIG. 5  makes it highly suitable for integration in a single integrated circuit or multi-chip module. 
     It will be apparent to the skilled person that many modifications can be made to the described embodiment without departing from the scope of the invention. For example, the I and Q baseband signals need not be differential signals, in which case alternative means must be provided for providing the mixers in each pair with antiphase baseband inputs. In another potential variant, the baluns are replaced with solid state subtractors.