Abstract:
Systems and methods for achieving high directivity (&gt;20 dB) coupling over a reasonable frequency bandwidth on a microstrip transmission line. An exemplary coupler cancels out-of-phase, coupled reflected power signals on the transmission line thereby increasing the directivity.

Description:
BACKGROUND OF THE INVENTION 
       [0001]    Standard RF/microwave couplers etched on microstrip have very poor directivity, typically ˜5 dB. Other modified microstrip couplers can achieve 20 dB directivity, but involve narrow etched line widths and spacings that require tight etching tolerances that may not be achievable or repeatable for low cost, high volume production. Also, these modified designs cannot be analyzed for proper function with standard linear simulators. They can only be analyzed with more sophisticated and expensive electromagnetic (EM) simulators. Without an EM simulator, a modified design with improved directivity is not possible in any kind of cost effective or timely manner. 
       SUMMARY OF THE INVENTION 
       [0002]    The present invention solves the problem of achieving high directivity (&gt;20 dB) coupling over a reasonable frequency bandwidth on a microstrip transmission line without the need for EM simulation, narrow line widths/spacings, or tight tolerances. The present invention can be implemented in any type of transmission line. It is especially suited to microstrip transmission lines. 
         [0003]    An exemplary coupler device includes a combiner, first and second coupling units connected between the combiner and a to-be-measured transmission line. The first and second coupling units comprise first and second coupling devices being in electrical communication with a to-be-measured transmission line, at least one first transmission line coupled between the combiner and the first coupling device and at least one second transmission line coupled between the combiner and the second coupling device. The at least one first and the at least one second transmission line have predefined impedance and phase delay values. The phase delay value of the at least one first transmission line differs from the phase delay value of the at least one second transmission line based on a phase delay value of the to-be-measured transmission line. 
         [0004]    In one aspect of the invention, the impedance of the at least one first transmission line is approximately equal to the impedance of the at least one second transmission line. 
         [0005]    In another aspect of the invention, the combiner has an isolation value generally greater than 20 dB. 
         [0006]    In still another aspect of the invention, each of the first and second coupling units includes a load resistor coupled between a node that is between an end of the first and second transmission lines and the respective coupling device and an electrical ground. The combiner has an isolation value generally less than 20 dB. 
         [0007]    In yet another aspect of the invention, the at least one first transmission line comprises first and second sub transmission lines and the at least one second transmission line comprises first and second sub transmission lines. The first sub transmission lines have first ends connected to the coupling device. Each of the first and second coupling units includes a load resistor coupled to second ends of the first sub transmission lines and first ends of the second sub transmission lines. Second ends of the second sub transmission lines are coupled to the coupling devices. Phase delay for at least one of the first or second sub transmission lines is equal. 
         [0008]    In still yet another aspect of the invention, the to-be-measured transmission line is located between a transmitter and an antenna. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0009]    Preferred and alternative embodiments of the present invention are described in detail below with reference to the following drawings: 
           [0010]      FIGS. 1-3  are schematic drawings showing different configurations formed in accordance with embodiments of the present invention; and 
           [0011]      FIG. 4  shows a transmission line with an equivalent in capacitors and an inductor. 
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0012]      FIG. 1  shows an exemplary microstrip coupler  20  that is capable of coupling power in a forward direction (P f ) on a transmission line Z 1 , while coupling very little reflected power (P r ) along the same transmission line Z 1 , thus achieving high directivity. 
         [0013]    In one embodiment, the coupler  20  is used to detect P f  along the microstrip transmission line Z 1  located between a transmitter  26  and an antenna  28 . The coupler  20  sends a sensed power value to a Power Detector Circuit  30 . 
         [0014]    The Power Detector Circuit  30  transforms the RF power to a voltage level that is proportional to the RF power level. The voltage is then sent to a field programmable gate array (FPGA) for processing. 
         [0015]    The coupler  20  includes a combiner  40  and a first coupler unit  42  and a second coupler unit  44 . Each coupler unit  42 ,  44  includes a coupling device (e.g., resistive, inductive or capacitive device) and a predefined lengths of transmission line Z 2 , Z 3 . The lengths depend on the type of combiner (i.e. in phase or quadrature type combiner). For example, resistive coupling is achieved with a chip or thin film resistor, capacitive coupling is achieved with a chip, printed or gap capacitor. The combiner  40  has reasonably high isolation (i.e. Wilkinson, branch line, rat race hybrid, or comparable combiner). Generally greater than 20 dB is considered a high isolation value. 
         [0016]    For the case of the combiner being a Wilkinson (in phase type combiner), let impedance for the microstrip transmission lines be as follows Z 1 =Z 2 =Z 3 =50 Ohm , and Z sh1  and Z sh2  have gap capacitance values of 0.029 pF, an approximate 37 dB coupling is achieved. Also let the phase delays for the respective microstrip transmission lines be as follows θ 1 =90°, θ 2 =90°, and θ 3 =0° at a particular frequency f o , f o  is the expected frequency of the transmitted signal. 
         [0017]    Forward power enters Port  1  and exits at Port  2 . A small amount of forward power P f  is coupled off from Z sh1 , travels thru Z 2  and is incident on the combiner at −90°. Forward power P f  travels thru Z 1  and a small amount of P f  is coupled off from Z sh2 , travels thru Z 3  and is incident on the combiner at −90°. The two coupled signals from forward power P f  are incident on the combiner  40  in phase and thus are added. 
         [0018]    The reflected (or reverse) power P r  enters Port  2  and exits at Port  1 . A small amount of reflected power P r  is coupled off from Z sh2 , travels thru Z 3  and is incident on the combiner at 0°. Reflected power travels thru Z 1  and a small amount is coupled off from Z sh1 , travels thru Z 2  and is incident on the combiner at −180°. The two coupled signals from reverse power P r  are incident on the combiner  40  180° out of phase and thus are canceled. 
         [0019]    Directivity is defined as forward coupled power minus reflected coupled power, typically expressed in dB. Theoretical analysis indicates directivity to be ≧20 dB for a bandwidth of about 19% for the above values of Z 1 , Z 2 , Z 3 , Z sh1  and Z sh2  when using a Wilkinson combiner. 
         [0020]    Different values of phasing for θ 1 , θ 2  and θ 3  will be required when using a branch line, rat race or other hybrid as the combiner as one of ordinary skill would be able to determine. Different values for Z 1 , Z 2 , Z 3 , Z sh1  and Z sh2  will result in different coupling, directivity and bandwidths. The values can be different, but typically Z 1 =Z 2 =Z 3  and Zsh 1 =Zsh 2 . 
         [0021]      FIG. 2  illustrates a coupler  80  with a combiner  82  that has lower isolation (i.e. broadband resistive “star” or “tee”). Operation of the coupler  80  is basically the same as the coupler  20  shown in  FIG. 1 . Two load resistors  86 ,  88  improve the directivity when the isolation of the combiner  82  is lower than 20 dB. As an example, when using a broadband resistive “star” combiner (isolation ˜6 dB), the directivity of the coupler  80  is ˜6.3 dB without load resistors  86 ,  88 , and &gt;20 dB with load resistors  86 ,  88 . 
         [0022]      FIG. 3  illustrates a coupler  90  having a combiner  92  that has lower isolation (i.e. broadband resistive “star” or “tee”). The coupler  90  includes load resistors  96 ,  98  that are placed between first microstrip transmission lines  100 ,  102  and second microstrip transmission lines  104 ,  108 . This is different than the coupler  80  shown in  FIG. 2 ; the ground on the resistors have been replaced with λ/4 transmission lines  100 ,  102  that have the same phase delay  110 ,  112 )(˜90°). λ is the expected wavelength of the received signal. λ/4 transmission line transforms an open circuit to a short circuit, thereby creating a virtual ground. Zsh 1  and Zsh 2  have extremely high impedance, almost an open circuit. This extremely high impedance transforms to an extremely low impedance through the λ/4 transmission lines  100 ,  102 . 
         [0023]    The coupler includes a second set of microstrip transmission lines  104 ,  108  with respective phase delay  114 ,  116  that is equal to the transmission lines Z 2 , Z 3  shown in  FIG. 2 . Phase delay of sub transmission lines  100 ,  102  are equal and generally 90 degrees. Phase delay of transmission lines  104 ,  108  are not necessarily equal. 
         [0024]      FIG. 4  shows that a transmission line, like the ones described above, can be replaced by other circuit components and still provide the same capabilities. A transmission line  120  is an etched trace on a circuit board with a specific width and length that achieves 50 Ohm and 90 degrees phase delay. A lumped element circuit  124  is electrically equivalent at a frequency of 1 GHz for the values given. Thus, in particular for lower frequency applications, a lumped element circuit or other transmission line equivalent could replace the transmission lines described above. 
         [0025]    While the preferred embodiment of the invention has been illustrated and described, as noted above, many changes can be made without departing from the spirit and scope of the invention. Accordingly, the scope of the invention is not limited by the disclosure of the preferred embodiment. Instead, the invention should be determined entirely by reference to the claims that follow.