Abstract:
An electronic drive for vector control of an induction motor controls slip and operating frequency in response to changes in stator voltage. The drive includes a torque control loop, a flux control loop and a frequency control loop. The control is based on a commanded stator current that is resolved into a torque-producing, or q-axis, current component and a flux-producing, or d-axis, current component that are in quadrature. The frequency control loop includes slip control in which a slip frequency command is produces based on a value for the leakage inductance of the motor. The leakage inductance value dynamically varies as a function of the q-axis current reference command.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
   Not Applicable 
   STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT 
   Not Applicable 
   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to high performance, electronic motor drives for variable speed or torque control of AC induction motors, and more particularly, to such motor drives which use vector control techniques and velocity feedback. 
   2. Description of the Related Art 
   Motor drives are commonly employed to control the application of electricity to a three-phase AC induction motor. Such motor drives include an inverter which switches DC voltage to output lines in a pulse width modulated (PWM) manner to control the frequency and amount of voltage applied to the motor and thus the motor velocity. 
   Vector control or field-oriented control is one technique used in a motor drive to control the speed and torque of the motor. With this technique, stator current is resolved into a torque-producing, or q-axis, current component I qs  and a flux-producing, or d-axis, current component I ds , where the q-axis current component leads the d-axis component by a 90° phase angle. This type of motor drive also requires knowledge of several motor parameters, such as inductance and resistance of the rotor and stator coils. 
   For accurate control of a three-phase motor, besides controlling the stator current frequency, it is also necessary to effectively control the slip, which represents the difference between the frequency of the stator current and the electrical frequency of the rotor rotation speed. The slip control is a key component of the high performance motor control to establish an accurate torque control. 
   U.S. Pat. No. 5,032,771 describes a high performance motor drive which controls the torque, frequency and slip at which the motor operates. The drive includes a torque control loop, a flux control loop, and a frequency control loop that incorporates slip management in response to a voltage difference. The slip is controlled in response to an error between a d-axis reference voltage and a d-axis feedback voltage. Flux weakening is provided in response to an error between a q-axis reference voltage that is sensed when the motor is operating at the base speed and a q-axis feedback voltage that is sensed when the motor is operating above the base speed. 
   Accurate slip control requires precise information about leakage inductance of the motor. The prior motor drives controlled slip based on an assumed constant value for the leakage inductance. However, the leakage inductance varies due to saturation effects as the motor load increases. Therefore, accurate torque control becomes difficult over a wide torque range when a constant value for the leakage inductance is used for slip control. 
   Therefore, it is desirable to provide an improved motor control technique that addresses the effects resulting from variation of the leakage inductance. 
   SUMMARY OF THE INVENTION 
   A method for controlling slip in an induction motor that has a stator and a rotor, comprises determining voltage feedback that is representative of actual stator voltage and determining a rotor frequency which is related to the rotational speed of the rotor. A current command is generated in response to the rotor frequency, the voltage feedback, and a desired velocity command. 
   A leakage inductance value is derived as a function of the current command and thus varies with changes in that command. The leakage inductance value is employed to produce a slip frequency command which in turn is used along with the rotor frequency to determine stator operating frequency command. The actual current flowing through the stator is measured and the resultant measurement is employed to determine a current feedback. The voltage applied to the stator is controlled in response to the stator operating frequency command, the current command and the current feedback. 
   Therefore unlike previous motor drives, the present method adjusts the value of the leakage inductance which is used in deriving the voltage commands that control the motor. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a schematic block diagram of a motor drive according to the present invention; 
       FIG. 2  is a detailed diagram of the slip regulator in  FIG. 1 ; and 
       FIG. 3  is a graph depicting a relationship between the leakage inductance and the q-axis current reference command produced in the motor drive. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1  illustrates a current-regulated, pulse width modulation motor controller, also called a “motor drive”,  10  for an alternating current (AC) induction motor  12 . The motor drive  10  includes a power section that receives electricity from a three-phase power supply  14 . The three phases are connected to an alternating current (AC) to direct current (DC) power converter  16  that rectifies the alternating currents from the power supply  14  to produce a DC voltage on a bus  15 . The DC bus  15  is connected to a pulse width modulation (PWM) voltage inverter  18 , which completes the power section of the motor drive  10 . The AC-DC power converter  16  also produces a feedback control signal V BUS  that indicates the voltage level on the DC bus  15 . 
   The conventional PWM voltage inverter  18  includes a group of solid state switching devices which are turned on and off by control signals to convert the input DC voltage to pulses of constant magnitude on three output lines connected to the motor  12 . The pattern of pulses on each output line from the PWM voltage inverter  18  is characterized by a first set of positive-going pulses of equal magnitude but of varying pulse width, followed by a second set of negative-going pulses of equal magnitude and varying pulse width. The rms value of this voltage pulse pattern approximates one cycle of a sinusoidal AC waveform. The pattern is repeated to generate subsequent cycles of that waveform. 
   To control the frequency and magnitude of the resultant AC power signals, the PWM voltage inverter  18  receives three balanced control signals, V as *, V bs * and V cs * which vary in phase by 120°. The magnitude and frequency of these control signals respectively determine the widths and frequency of the pulses in the three power signals which are applied to the terminals of the motor  12 . 
   As used throughout this description, an asterisk associated with a signal designation denotes a “command” signal and a designation without an asterisk denotes a signal applied to derived from signals applied to the motor  12 . An “s” subscript in a signal designation indicates that the associated signal is referred to the motor stator. 
   The AC inverter input control signals, V as *, V bs * and V cs * result from a phase conversion which is accomplished with a 2-to-3 phase converter  20 , which includes a synchronous to stator converter at its inputs. The input signals V qs   e * and V ds   e * to that phase converter are synchronous voltage command signals of a given steady state magnitude. These signals are related to a stationary d-q reference frame in which torque-controlling electrical parameters are related to a q-axis and flux-controlling electrical parameters are related to a d-axis. The q-axis signal leads the d-axis signal by 90° in phase. The voltage commands V qs   e * and V ds   e * are produced by a synchronous current regulator  26  in response to measurements of the phase currents I as , I bs  and I cs  flowing through the stator terminals on the motor  12 , along with other input parameters, as will be described. 
   The motor phase currents I as , I bs  and I cs  are measured by conventional current sensing devices  22 . A first 3-to-2 phase converter  24 , which also includes a stator to synchronous converter, transforms these phase current measurements into current feedback signals I qs   eFB  and I ds   eFB  related to the synchronous d-q frame of reference. The stator terminal voltages V as , V bs and V   cs  are applied to inputs of a second 3-to-2 phase converter  27  which transforms those phase voltages to feedback signals V qs   FB  and V ds   FB  which also are related to the synchronous d-q frame of reference. A conventional encoder  28  is attached to the shaft of the motor  12  and produces a signal indicating the angular position θ r  of that shaft. This encoder signal is applied to a position-to-velocity conversion circuit  30  which generates a digital rotor velocity signal ω r . That velocity signal ω r  is combined with an angular slip frequency command ω s * at first summation node  31  to produce a stator operating frequency command ω e * which is fed to the synchronous current regulator  26 . Generation of the slip frequency command ω s * will be described hereinafter as part of the description of the slip controller  34 . 
   The synchronous voltage commands V qs   e * and V ds   e * are produced by the synchronous current regulator  26  which includes a proportional-integral (PI) control loop with summing inputs. A q-axis current reference command I qs   e *, received at one input, is algebraically summed with the current feedback signal I qs   eFB  to provide a current error for the q-axis. A d-axis current reference command I ds   e * at another input is algebraically summed with the I ds   eFB  current feedback signal to provide a current error for the d-axis. The q-axis and d-axis current reference commands are collectively referred to herein as a current command. The synchronous current regulator  26  employs these input signals to produce the voltage reference commands V qs   e * and V ds   e * based on the current errors. 
   The d and q axis current reference commands I ds   e * and I qs   e * are supplied to synchronous current regulator  26  by a field-oriented controller  32  and a slip controller  34 , both of which can be implemented by a commercially available microcontroller that operates in response to a stored program. The field-oriented controller  32  is described in detail in U.S. Pat. No. 5,032,771, the description of which is incorporated herein by reference. The motor drive  10  receives a desired velocity command ω r * as an input, which the field-oriented controller  32  employs in to furnishing digital values for the torque related q-axis current reference command I qs   e * and the flux related d-axis current reference command I ds   e * to the synchronous current regulator  26 . The present invention can be used with other types of field-oriented controllers. Alternatively the motor drive may receive a desired torque command instead of the desired velocity command. The present invention can be used with other types of field-oriented controllers. 
   The slip controller  34  includes a flux regulator  36  which receives the motor voltage feedback signals V qs   FB  and V ds   FB  from the second 3-to-2 phase converter  27  and the feedback signal V BUS  which indicates the voltage level on the DC bus  15 . In response to those input signals, the flux regulator  36  generates the d-axis current reference command I ds   e * in the synchronous d-q frame of reference, as described in the U.S. patent mentioned immediately above. The d-axis current reference command is applied as an input to the field-oriented controller  32 . 
   The slip controller  34  incorporates a novel slip regulator  38 , the details of which are shown in  FIG. 2 . The slip regulator  38  receives the current reference commands I qs   e * and I ds   e *. The q-axis current reference command I qs   e * is applied to function  40  which calculates the d-axis voltage command V ds   e * according to the equation:
 
 V   ds   e *=( r   s   I   ds   e *)−(ω e (σ L   s ) I   qs   e *)  (1)
 
where r s  is the stator resistance, ω e * is the stator operating frequency command, and σL s  is the leakage inductance. The leakage inductance in turn is defined by the expression:
 
                   σ   ⁢           ⁢     L   s       =       L   s     -       L   m   2       L   r                 (   2   )               
where L s  is the inductance of the stator, L m  is the magnetizing inductance, and L r  is the inductance of the rotor.
 
   Computation of the d-axis voltage reference command V ds   e * commences at a first multiplier  42  where the q-axis current reference command I qs   e * is multiplied by the leakage inductance σL s . The leakage inductance is provided by a look-up table  44  based on the magnitude of that q-axis current reference command. As noted previously the leakage inductance varies due to saturation effects as the motor load increases.  FIG. 3  graphically depicts the contents of the look-up table  44  and illustrates the relationship between the q-axis current reference command and the leakage inductance. This relationship is determined empirically for the specific motor  12  that is connected to the motor drive  10 . This determination can be performed automatically by the motor drive during commissioning as described in U.S. Pat. No. 5,689,169. The data gathered during this process is stored in the memory of the motor drive as the look-up table  44 . During operation of the motor thereafter the value of the q-axis current reference command I qs   e * addresses the storage location in the look-up table that contains the related value of the leakage inductance σL s . Therefore, unlike prior drives which used a constant value for the leakage inductance, the present slip regulator  38  uses a leakage inductance value σL s  that changes in correspondence with the actual variation of the leakage inductance of the motor. 
   The output produced by the first multiplier  42  is applied to one input of a second multiplier  46  which also received the stator operating frequency command ω e *. The product from the second multiplier  46  is applied to an inverting input of a second summation node  48 . The d-axis current reference command I ds   e * is multiplied by a constant value for the stator resistance r s  by a third multiplier  50  and the product is applied to a non-inverting input of the second summation node  48 . The stator resistance r s  of the particular motor  12  is measured during the commissioning of the motor drive  10  and stored in the drive&#39;s memory. The second summation node  48  produces the d-axis voltage command V ds   e * from which the motor voltage feedback signal V ds   FB  is subtracted at a third summation node  52  to generate a voltage error signal V ERR . 
   Function block  54  changes the polarity of the voltage error signal V ERR  if the product of the q-axis current reference command I qs   e * and the stator operating frequency command ω e * is a negative value. The resultant error value then is applied to a proportional-integral control loop  55  the comprises an integral branch  56  and a proportional branch  58  which produces a value for a slip gain K s  according to the expression:
 
 K   S   =K   i   ∫[V   ds   e   *−V   ds   *]+K   PS   [V   ds   e   *−V   ds *]  (3)
 
The integral branch  56  provides the first term of that expression as designated by the integral function 1/S, where K i  is a constant multiplication factor for the integral. In the proportional branch  58  the error value from function block  54  is multiplied by a proportional constant K PS . The values produced by the two proportional-integral control branches  56  and  58  are summed at node  60  to produce the slip gain K s  that then is multiplied by the q-axis current reference command I qs   e * in a third multiplier  62  to produce the slip frequency command ω s * at the output of the slip regulator  38  wherein:
 
ω s   *=K   s ( I   qs   e *)  (4)
 
   Referring again to  FIG. 1 , the slip frequency command ω s * is summed with the rotor frequency feedback ω r  at the first summation node  31  to generate the stator operating frequency command ω e *. This value is fed back to slip regulator  38 . Therefore, the current regulator  26  produces the voltage reference commands V qs   e * and V ds   e *. 
   The slip frequency command ω s * also is integrated at operation  66  to obtain a desired angular slip position θ s  which is arithmetically summed with the rotor angular position θ r  to derive an angular position of the stator magnetomotive force θ e . The stator magnetomotive force position is used by the various phase converters  20 ,  24  and  27  of the motor drive  10 . 
   The foregoing description was primarily directed to preferred embodiments of the present invention. Although some attention was given to various alternatives within the scope of the invention, it is anticipated that one skilled in the art will likely realize additional alternatives that are now apparent from disclosure of embodiments of the invention. Accordingly, the scope of the invention should be determined from the following claims and not limited by the above disclosure.