Abstract:
A circuit arrangement and control thereof for igniting a high intensity discharge (HID) lamp, for reducing the variation of the resonant ignition voltage under the parasitic capacitive loading condition, and for increased circuit stability. The high frequency ignition voltage is only applied to the lamp during an ignition phase. The variation of the magnitude of the resonant ignition voltage with respect to the parasitic capacitance at the lamp leads is minimized by inserting a damping resistor in series with the ignition resonant capacitor. In a normal operation after ignition, the charge and/or discharge current of the ignition resonant capacitor is bypassed through a bypass device instead of flowing through a current sense resistor, so that only a chopper current flows through a sensor by paralleling a relatively high impedance resistor with the sensor.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    The present invention pertains to an apparatus and method for driving a high intensity discharge (HID) lamp. Specifically, the present invention is directed to generating a high frequency resonant ignition voltage to ignite (start) the HID lamp, and to maintain a stable circuit operation with minimal interference from a high frequency resonant ignition circuit to a peak current sense signal that is used for output power and current control during a normal state of operation. A variation of the magnitude of the resonant ignition voltage with respect to a parasitic capacitance related to a length of the lamp leads is minimized by the inclusion of a damping resistor connected in series with a resonant ignition capacitor.  
           [0003]    2. Discussion of Background and Relevant Information  
           [0004]    Electronic high intensity discharge lamps generally employ one of two techniques for igniting (starting) the lamp. In a first technique, the lamp is ignited using a pulsed ignition method. In a second technique, the lamp is ignited using a resonant ignition method. The peak magnitude of the ignition voltage associated with the resonant ignition method is lower than the peak magnitude of the ignition voltage associated with the pulse ignition method. Accordingly, from the standpoint of safety, the resonant ignition method is generally preferred over the pulsed ignition method.  
           [0005]    Further, two distinctively different methods exist to continue operating the lamp after it has been ignited. In a first method, the lamp is operated with a high frequency signal that is typically in the kilo Hertz (kHz) range. In a second method, the lamp is operated with a low frequency signal that is typically measured in the hundreds of Hertz range. Due to acoustic resonance problems associated with high frequency operations, it is generally preferred to employ the low frequency operation method to maintain the operation (e.g., illumination) of the lamp.  
           [0006]    In order to generate a high frequency voltage having sufficient energy to ignite the lamp or to run the lamp (after ignition) with a high frequency signal, three fundamental approaches are generally taken, as shown in FIGS.  3 ( a ) to  3 ( c ).  
           [0007]    [0007]FIG. 3( a ) illustrates a discharge lamp driving circuit having a chopper and a high frequency inverter. Depending upon different control schemes applied to switches Q 1  to Q 4 , this configuration can serve many design purposes.  
           [0008]    It is known that HID lamps exhibit an acoustic resonance when operated at a high frequency. U.S. Pat. No. 4,912,374 discloses a method to interrupt the high frequency current with a smoothed DC current. Inductor L 1  and capacitor C 1  form a buck resonant network. Transformer T and capacitor C 2  form an inverting resonant circuit. When transistor pair Q 1  and Q 4  and transistor pair Q 2  and Q 3  are alternately switched at a high frequency, two high frequency AC currents flow through the lamp. The first high frequency AC current is produced by the buck resonant network. The second high frequency AC current is produced by the inverting resonant network. As a result, a loop current is formed between the capacitor C 1 , the transformer T, and the lamp. When transistor Q 4  is switched at a high frequency, transistor Q 1  is ON, and transistors Q 2  and Q 3  are completely OFF (due to the chopper, or buck, configuration.), so that a DC current flows from left to right through the lamp. When transistor Q 3  is switched at a high frequency, transistor Q 2  is ON, and transistors Q 1  and Q 4  are completely OFF, so that a DC current flows from right to left through the lamp.  
           [0009]    To control the DC current, some sort of buck current sensing is required. Such a system is not disclosed in detail in U.S. Pat. No. 4,912,374. The simplest method to sense the buck current is to add a sense resistor in series with input bus voltage V 1 . However, unless special care is taken to separate the inverting resonant network current from the buck network current, a coupling may occur between the inverting resonant network and the buck resonant network. U.S. Pat. No. 4,912,374 does not disclose the separation of the inverting high frequency operation and the buck DC or low frequency operation, but the inverting high frequency operation is utilized just for starting (igniting) the lamp and the DC (or low frequency) operation is utilized for the normal (continuous) operation of the lamp after it has been started.  
           [0010]    [0010]FIG. 3( b ) illustrates a modification of U.S. Pat. No. 4,912,374, in which MOSFET Q 5  and diode D 5  are added. The inclusion of these components results in the lamp current comprising a clean square wave, while the sensed buck current comprises a clean triangular wave. It is noted that MOSFET Q 5  can be switched OFF any time after the lamp is ignited (started), or whenever the high frequency current is not needed for the lamp operation. When MOSFET Q 5  is switched OFF, the buck network, formed by inductor L 1  and capacitor C 1 , and the ignition network, formed by transformer T and ignition capacitor C 2 , are completely decoupled. That is, ignition capacitor C 2  is electrically disconnected from the circuit. There is no charge (or discharge) current flowing through the ignition capacitor C 2  or the current sensing resistor Rs, due to the switching of transistors Q 1  and Q 2 . Further, diode D 5  prevents any voltage overshoot during the switching of MOSFET Q 5 .  
           [0011]    The disadvantage of this modification is that a high voltage MOSFET Q 5  and a high voltage diode D 5  is required, along with any associated driving circuitry required to drive MOSFET Q 5 . This increases the circuit complexity and increases the manufacturing costs. It is noted that if the composite waveform of the high frequency current and the DC current is required to prevent an acoustic resonance, MOSFET Q 5  has to be turned ON during the high frequency period and turned OFF during the low frequency period.  
           [0012]    A dual stage output filter of U.S. Pat. No. 6,020,691 is illustrated in FIG. 3( c ), in which a chopper (or buck) power regulator with a high frequency resonant ignition, a discontinuous first resonant stage inductor current, and a continuous second resonant stage inductor current are related to each other.  
           [0013]    In U.S. Pat. No. 6,020,691, a first stage resonant frequency fr 1 , formed by inductor L 1  and capacitor C 1 , is lower than a second stage resonant frequency fr 2 , formed by inductor L 2  and capacitor C 2 . In addition, a distance between the first stage resonant frequency fr 1  and the second stage resonant frequency fr 2  is somewhat confined not to be less than a selected minimum value, in order to avoid an excessive resonant current circulating in the circuit. The ignition voltage is generated by sweeping the frequency over the second stage resonant frequency, fr 2 . For example, if the second stage resonant frequency fr 2  is selected to be, for example, approximately 40 kHz and a minimum sweeping frequency is selected to be, for example, approximately 30 kHz, the first stage resonant frequency fr 1  may be selected to be, for example, approximately 22 kHz. This kind of circuit arrangement suffers from frequency inaccuracies and component tolerance problems, because the circulating current of the first stage resonant network is highly related to the frequency fr 1  and the minimum sweeping frequency. A further disadvantage of this circuit arrangement is that the magnitude of the ignition pulse, which is mainly generated by the second stage network, is a function of both resonant frequencies, since two stages are cascaded together. The input voltage signal, with its frequency near the second stage resonant frequency, is damped by the first stage network and amplified by the second stage network. Thus, the Q factor of the second stage network has to be significantly high so that enough ignition voltage can be generated.  
         SUMMARY OF THE INVENTION  
         [0014]    The present invention overcomes the inability of the prior art to electrically separate (isolate) the first resonant network design and the second resonant network design. According to the present invention, the ignition capacitor is isolated from the circuit to prevent a charge current (and/or discharge current) from interfering with a load current sense circuit.  
           [0015]    According to a feature of the invention, a relatively “clean” signal is provided to a current sensing circuit of a buck regulator, even when a relatively high spike current is fed to the ignition capacitor.  
           [0016]    According to an advantage of the present invention, a damping device, such as, for example, a damping resistor, is provided, such that a variation of a peak ignition voltage that is generated is limited to a minimal parasitic capacitance, such as, for example, a few hundred pico-farads, at the output.  
           [0017]    According to another object of the invention, leakage current through the path of a bypass diode is significantly less than the current flowing through the sensing resistor, so that the current sensing is not affected by the diode leakage current.  
           [0018]    According to an object of the present invention, a discharge lamp driving circuit, comprises a tank circuit, and a DC-AC inverter. The tank circuit has two resonant networks, and a lamp driving connection. The lamp driving connection is electrically connected to a lamp. The DC-AC inverter is electrically connected to a voltage input and to the tank circuit. A first resonant network of the tank circuit delivers an alternating rectangular current during a normal operation mode, while a second resonant network of the tank circuit delivers an alternating resonant ignition voltage during a starting operation mode. The tank circuit is configured so that the second resonant network includes at least one damping resistor in series with at least one resonant capacitor.  
           [0019]    According to a feature of the invention, the DC-AC inverter may be either a full bridge inverter or a half bridge inverter.  
           [0020]    According to an advantage of the invention, the first resonant network comprises a capacitor and an inductor that are electrically connected in series, and the first resonant network is connected to an output of the bridge circuit.  
           [0021]    According to another advantage of the invention, the second resonant network comprises an inductor, a capacitor and a damping device. The inductor, capacitor and damping device are electrically connected in series, with the second resonant network being connected to an output of the bridge inverter and the bypass device.  
           [0022]    A further advantage of the invention resides in the lamp driving connection being electrically connected in series with an inductor of the second resonant network and an inductor of the first resonant network, with the serially connected lamp driving connection and inductor of the second resonant network being further connected in parallel with a capacitor of the first resonant network.  
           [0023]    Another object of the present invention resides in a discharge lamp driving circuit that comprises a tank circuit and a DC-AC inverter. The tank circuit has a first resonant network, a second resonant network, and a lamp driving connection. The lamp driving connection is electrically connected to a lamp. The DC-AC inverter, which is electrically connected to a voltage input and to the tank circuit, includes a sensing device, a bypass device associated with the second resonant network and a bridge circuit. The sensing device operates to sense an amount of current of the first resonant network in the tank circuit, while the bypass device operates to decouple the current flow of the second resonant network from the current flow of the first resonant network.  
           [0024]    According to an advantage of the invention, the sensing device, which may be, for example, a sensing resistor, is connected in parallel with the bypass device.  
           [0025]    According to a feature of the invention, the DC-AC inverter includes a bridge inverter, such as, for example, a full bridge inverter or a half bridge inverter.  
           [0026]    A still further feature of the invention resides in the first resonant network comprising a capacitor and an inductor that are electrically connected in series, the first resonant network being connected to an output of the bridge inverter.  
           [0027]    Another feature of the invention pertains to the second resonant network comprising an inductor, a capacitor and a damping device. The inductor, capacitor, and damping device are electrically connected in series, the second resonant network being connected to an output of the bridge inverter and the bypass device.  
           [0028]    It is noted that the lamp driving connection may be electrically connected in series with an inductor of the second resonant network and an inductor of the first resonant network. The serially connected lamp driving connection and inductor of the second resonant network being further connected in parallel with a capacitor of the first resonant network.  
           [0029]    Further, the sensing device may be connected between a first input connection of the voltage input and a first input of a bridge inverter, a second input of the bridge inverter being connected to a second input connection of the voltage input.  
           [0030]    According to another object of the invention, a discharge lamp driving circuit comprises a tank circuit and a DC-AC inverter. The tank circuit includes a first resonant network, a second resonant network, and a lamp driving connection, with the lamp driving connection being electrically connected to a lamp. The DCAC inverter is connected to a voltage input and the tank circuit. The first resonant network delivers an alternating rectangular current (which may having an operating frequency of less than approximately 1 kHz) to the lamp during a normal operation mode, while the second resonant network delivers an alternating resonant ignition voltage (which may have an operating frequency greater than approximately 20 kHz) to the lamp during a starting operation mode. Additionally, the second resonant network includes at least one damping resistor in series with at least one resonant capacitor. The DC-AC inverter includes a sensing device, a bypass device, and a bridge inverter. The sensing device senses a current flow of the first resonant network while the bypass device de-couples a current flow of the first resonant network and a current flow of the second resonant network.  
           [0031]    According to a feature of the invention, the sensing device is connected in parallel with the bypass device, which may comprise, for example, two series connected diodes. The sensing device may be connected between a first input connection of the DC voltage input and a first input of a bridge inverter, while a second input of the bridge inverter is connected to a second input connection of the DC voltage input.  
           [0032]    A junction of the two series connected diodes may be connected to the second resonant network. In addition, a leakage preventing device may be connected in series with at least one of the two series connected diodes. The leakage preventing device exhibits a resistance that is greater than a resistance of the sensing device. Preferably, the resistance of the leakage preventing device is equal to at least twenty times said resistance of the sensing device.  
           [0033]    According to another feature of the invention, the DC-AC inverter may be a full bridge inverter or a half bridge inverter.  
           [0034]    According to an advantage of the invention, the first resonant network may comprise a capacitor and an inductor that are electrically connected in series, with the first resonant network being connected to an output of the bridge inverter. Further, the second resonant network may comprise an inductor, a capacitor and a damping device, in which the inductor, the capacitor and the damping device are electrically connected in series, the second resonant network being connected to an output of the bridge inverter and the bypass device.  
           [0035]    The lamp driving connection may be electrically connected in series with an inductor of the second resonant network and an inductor of the first resonant network, with the serially connected lamp driving connection and inductor of the second resonant network being further connected in parallel with a capacitor of the first resonant network.  
           [0036]    According to a still further object of the invention, a method is disclosed for driving a discharge lamp. According to the method, a lamp is electrically connected to a lamp driving output of a tank circuit having a first resonant network and a second resonant network, with a voltage being input to the tank circuit from a DC-AC inverter. The DC-AC inverter is electrically connected to a voltage input. The DC-AC inverter includes a sensor, a bypass device associated with the second resonant network, and a bridge circuit. An amount of current flowing in the first resonant network of the tank circuit is monitored with the sensor, while the current flowing in the second resonant network is decoupled from a current flowing in the first resonant network with the bypass device.  
           [0037]    According to a feature of the invention, the sensor may be connected in parallel with the bypass device. The bypass device may comprise two series connected diodes. In such a configuration, a leakage prevention device may be electrically connected in series with at least one of the two series connected diodes. The leakage prevention device preferably exhibits a resistance that is at least twenty times greater than a resistance of the sensing device.  
           [0038]    An additional feature of the invention is the inclusion of a bridge inverter, the first resonant network including a capacitor and an inductor that are electrically connected in series, the first resonant network being connected to the output of the bridge inverter.  
           [0039]    A still further feature of the invention is that the second resonant network includes an inductor, a capacitor and a damping device. The inductor, the capacitor, and the damping device are electrically connected to an output of the bridge inverter and the bypass device.  
           [0040]    A still further object of the invention pertains to a method for driving a discharge lamp that is electrically connected to a lamp driving output of a tank circuit, in which the tank circuit has a first resonant network and a second resonant network, the voltage being input to the tank circuit from the DC-AC inverter, the second resonant network having at least one damping resistor in series with at least one resonant capacitor, and in which a DC-AC inverter, having a sensor, a bypass device, and a bridge circuit is electrically connected to the DC voltage input and to the tank circuit. The method comprises operating the tank circuit such that the second resonant network delivers an alternating current ignition voltage to the lamp during a startup operation mode. After a lapse of a predetermined time period, the tank circuit is operated such that the first resonant network delivers an alternating rectangular current to the lamp during a normal operation mode. A current flow in the first resonant network is sensed with the sensor, while a current flow in the second resonant network is decoupled from the current flow in the first resonant network with the bypass device of the DC-AC inverter. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0041]    The foregoing and other objects, features and advantages of the invention will be apparent from the following more particular description of the preferred embodiment, as illustrated in the accompanying drawings, in which like characters refer to the same parts throughout the various views, and wherein:  
         [0042]    [0042]FIG. 1( a ) illustrates a block diagram of a lamp driving circuit of the present invention;  
         [0043]    [0043]FIG. 1( b ) illustrates a schematic diagram of an embodiment of the invention shown in FIG. 1( a );  
         [0044]    [0044]FIG. 1( c ) illustrates a high impedance path connection of the circuit shown in FIG. 1( b );  
         [0045]    FIGS.  2 ( a ),  2 ( b ), and  2 ( c ) illustrate various inductive elements that are utilized with the invention of FIG. 1( b );  
         [0046]    [0046]FIG. 3( a ) illustrates a prior art block diagram of an output network having an ignition resonant network that is separated from a buck resonant network;  
         [0047]    [0047]FIG. 3( b ) illustrates a modification of the circuit of FIG. 3( a ), in which the ignition resonant network is separated from the buck resonant network, with the ignition capacitor being switched OFF after the lamp is ignited;  
         [0048]    [0048]FIG. 3( c ) illustrates a prior art two stage LC output filter;  
         [0049]    [0049]FIG. 4 represents the circuit of FIG. 1( a ) with the inductive element of FIG. 2( b );  
         [0050]    FIGS.  5 ( a ) to  5 ( e ) show various ideal waveforms produced at predetermined points of the circuit shown in FIG. 4;  
         [0051]    [0051]FIG. 6( a ) illustrates the amount of current flowing through an ignition capacitor C 2  at a predetermined load voltage, a current flowing through a bypass diode D 2 , and a sensing voltage across a sensing resistor Rs, the electronic components being utilized in the circuit of FIG. 4;  
         [0052]    [0052]FIG. 6( b ) illustrates the amount of current flowing through the ignition capacitor C 2 , and the voltage on the sense resistor Rs without the bypass diode;  
         [0053]    [0053]FIG. 7( a ) illustrates an equivalent circuit for preventing leakage current through bypass diodes;  
         [0054]    FIGS.  7 ( b ) and  7 ( c ) illustrate associated waveforms of the circuit of FIG. 7( a );  
         [0055]    [0055]FIG. 8( a ) illustrates an ignition voltage envelope at the lamp ends with no leads; and  
         [0056]    [0056]FIG. 8( b ) illustrates the ignition voltage envelope at the lamp ends with  15  feet of lamp leads. 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0057]    [0057]FIG. 1( a ) illustrates a block diagram of the present invention. As shown in FIG. 1( a ), a driving circuit  5  comprises a DC-AC inverter  8 , and a tank circuit  12 . The tank circuit  12  operates (e.g., provides electrical power to) a lamp, such as, for example, a high intensity discharge lamp LAMP.  
         [0058]    The DC-AC inverter  8  comprises a DC-AC circuit  10 , a sensing device  14  and a bypass device  16 . The DC-AC inverter  8  may be formed as either a fill bridge inverter or a half bridge inverter.  
         [0059]    The tank circuit  12  comprises a first resonant network  18  and a second resonant network  20 .  
         [0060]    [0060]FIG. 1( b ) illustrates a specific circuit arrangement of the present invention. However, it is understood that variations therein may be made without departing from the spirit and/or scope of the instant invention.  
         [0061]    The bridge circuit  10  comprises a plurality of MOSFET transistors Q 1  to Q 4  that are configured in a full bridge arrangement. The first resonant network  18  of the tank circuit  12  comprises a capacitor C 1 , and an inductor L 1 . The second resonant network  20  of the tank circuit  12  comprises an ignition capacitor C 2 , a first resistor R 1 , and an inductive element T. The sensing device  16  comprises a sensing resistor Rs, while the bypass device comprises diodes D 1  and D 2  that are connected in series via a resistor R 2 .  
         [0062]    One wire lead of a high intensity discharge (HID) lamp LAMP is connected to a junction of capacitor C 1  and inductor L 1 , while a second wire lead of the HID lamp LAMP is connected to a junction B of the inductive element T. FIG. 1( b ) shows the electrical connection of the lamp with respect to the tank circuit  12 ; however, it is understood that this depiction of the lamp is for the purpose of conveying the electrical connection of the lamp to the first and second resonant circuits  18  and  20  of the tank circuit  12 , and is separate from (not a part of) the tank circuit  12 .  
         [0063]    Inductor L 1  and capacitor C 1  of the first resonant circuit  18  form a chopper (or buck) filter network. Inductive element T and ignition capacitor C 2  of the second resonant circuit  20  form a high frequency resonant network, while the resonant damping resistor R 1  functions to reduce a quality factor and widen a bandwidth of the high frequency resonant network.  
         [0064]    Bypass diodes D 1  and D 2  of the bypass device  16  control a current flow from the ignition capacitor C 2  through the sensing resistor Rs, while resistor R 2  operates to prevent (or reduce) any current from MOSFET Q 2  and/or MOSFET Q 4  from flowing through bypass diodes D 1  and D 2 . Preferably, resistor R 2  has a resistance value that is much higher (greater) than a resistance value of the sensing resistor Rs. Resistor R 2 , which is electrically installed in series with resistor R 1 , also functions as a damping resistor.  
         [0065]    According to a variation shown in FIG. 1( c ), resistor R 2  comprises a plurality (e.g., two in FIG. 1( c )) resistors, which may (or may not) be of equal value. In this variation, one resistor is connected in series with bypass diode D 2  while the other resistor is connected in series with bypass diode D 1 .  
         [0066]    During a starting (igniting) operation, an ignition network (formed by the inductive element T, capacitor C 2 , resistor R 1  and resistor R 2 ) is energized by a frequency varying and duty cycle varying source supplied to a voltage supply line Vbus, MOSFET Q 1 , and MOSFET Q 2 . By controlling the switching rate of MOSFETS Q 1  and Q 2  (e.g., the frequency at which the system turns ON and OFF), the frequency and/or duty cycle in each high frequency cycle can be linearly swept from a first (e.g., high) frequency, of, for example, approximately 200 kHz, to a second (e.g., low) frequency, of, for example, approximately 100 kHz.  
         [0067]    The following discussion is based on the assumption that the inductive element T shown in FIG. 2( b ) is used in the circuit of FIG. 1( b ). Given the above operating frequencies, first inductor T′ has a value of approximately 750 uH, capacitor C 2  has a value of approximately 1.5 nF, and any parasitic capacitance that may exist will be in the range of approximately 0 pF to a maximum of approximately 150 pF. The parasitic capacitance is reflected to the inductor T′ that forms part of a capacitive element of the resonant network with capacitor C 2 . Because of the reflection, the resonant frequency changes. Based upon simulations that have been performed, the resonant frequencies become equal to approximately 143 kHz; 135 kHz; 127 kHz; and 121 kHz in response to the presence of a parasitic capacitance equal to approximately 0 pF; 50 pF; 100 pF; and 150 pF, respectively.  
         [0068]    It is noted that the resonant frequency is not effected (changed) by the presence or absence of the damping resistor; however, variations in circuit gains with respect to the parasitic capacitance do occur. It is also noted that, variations in the circuit gains (with respect to the parasitic capacitance) is decreased when the damping resistor is included. For example, without the damping resistor, a peak gain decreases approximately 16.7 percent over a parasitic capacitance range from 0 pF to 150 pF. However, when a damping resistor R 1 , having a value of approximately 30 ohms is included in the circuit, the peak gain decreases only about 6.3 percent over the same 0 pF to 150 pF range.  
         [0069]    From the above, it is apparent that the inclusion of the damping resistor plays a role in stabilizing the gain. For a given driving source and frequency sweeping pattern, the peak magnitude of the ignition voltage is proportional to the gain of the resonant network. As noted above, the gain varies by only approximately 6.3 percent with the inclusion of the damping resistor, and thus, the ignition voltage stays within a predetermined limit when the parasitic capacitance varies from 0 pF to 150 pF. It is noted that in applications where the lamp leads may be up to approximately 15 feet, it was measured that the equivalent capacitance is approximately 150 pF.  
         [0070]    FIGS.  8 ( a ) and  8 ( b ) illustrate an ignition voltage envelope when the lamp leads are 0 feet and 15 feet in length, respectively. In this regard, it is noted that the Inductive Element T of FIG. 2( a ) has been employed, and further, that a small value capacitor (of, for example, approximately 150 pF) can be added across terminal points B and C (see FIG. 2( a )) to reduce (filter) some of the very high frequency ringing shown in the figures.  
         [0071]    During the starting (igniting) operation, MOSFET Q 3  and MOSFET Q 4  are switched in-phase with MOSFET Q 2  and MOSFET Q 1 , respectively. Since the resonant frequency of the chopper (buck) filter network (formed by inductor L 1  and capacitor C 1 ) is much lower than the sweeping frequencies of approximately 100 kHz to 200 kHz, only a small resonant voltage is generated across inductor L 1  or capacitor C 1 . Once the lamp is ignited during the starting operation, lamp current flows through MOSFET pair Q 1  and Q 4  at one-half of the high frequency cycle, and through MOSFET pair Q 2  and Q 3  at one-half of the high frequency cycle during the second half of the high frequency cycle.  
         [0072]    During the normal operation of the lamp (e.g., after the lamp is ignited/started), MOSFET Q 1  and MOSFET Q 2  operate at a low frequency of, for example, approximately 170 Hz, while MOSFET Q 3  and MOSFET Q 4  operate at a high frequency of, for example, approximately 50 kHz. Alternatively, MOSFET pair Q 1  and Q 4  may be operated at a high frequency of, for example, approximately 50 kHz, during a first half of a low frequency cycle of, for example, approximately 170 Hz, while MOSFET pair Q 2  and Q 3  are operated at the high frequency (e.g., approximately 50 kHz) during the other half of the low frequency cycle (e.g., approximately 170 Hz). During a normal operation with a low lamp voltage, all switches can operate in high frequency. In either case, the voltage at the junction of MOSFET Q 1  and MOSFET Q 2  (e.g., point A in FIG. 1( b )) is HIGH when MOSFET Q 1  is ON and LOW when MOSFET Q 2  is ON. During a high voltage to low voltage transition (or low voltage to high voltage transition), there is a charge (or discharge) of current flowing through the ignition capacitor C 2 .  
         [0073]    A detailed operation explanation for the case of a low lamp voltage will now be provided. FIG. 4 represents the circuit of FIG. 1( b ) with the Inductive Element T of FIG. 2( b ), while FIGS.  5 ( a ) to  5 ( e ) represent waveforms at various locations of the circuit of FIG. 4. The resonant frequency of the chopper (buck filter) network formed by capacitor C 1  and inductor L 1  is lower than the high frequency, of, for example, approximately 50 kHz, in normal operation. The resonant frequency of the inverter network (ignition network), formed by capacitor C 2 , resistor R 1 , and inductive element T′, is higher than the high frequency (e.g., approximately 50 kHz) during the normal operation. The voltage across capacitor C 1  may be considered to be constant during one high frequency cycle. The voltage across inductors T′ and T″ may be considered to be zero during one high frequency cycle.  
         [0074]    Lamp voltage V(lamp) is equal to the voltage on capacitor C 1 . Thus, the voltage at point C in FIG. 4 is equal to voltage V(1). At time t equals 0, MOSFET transistors Q 2  and Q 3  are ON and MOSFET transistors Q 1  and Q 4  are OFF. Chopper current V(Rs) ramps up and reaches a predetermined peak level, at which point, MOSFET Q 3  turns OFF and a freewheeling current starts to flow through inductor L 1 , capacitor C 1 , the lamp, and the internal diodes of MOSFET transistors Q 2  and Q 4 , until time t equals t 1 . At time t equals t 1 , MOSFET Q 2  turns OFF. A freewheeling current continues to flow through the internal diode of MOSFET Q 1  and back into the bus line, until the current reaches zero at time t equals t 2 . At time t equals t 2 , a new cycle initiates.  
         [0075]    At time t equals t 0  and time t equals t 1 , voltage V(1) suddenly switches from HIGH to LOW (or LOW to HIGH). The relationship between the voltage V(1) at point C and current I(C 2 ) flowing through capacitor C 2  can be expressed by the equation:  
           I ( C   2 )= C   2 * dV(1) / dt ,  
         [0076]    neglecting any parasitic inductance and the damping resistors R 1  and R 2 .  
         [0077]    The voltage across a capacitor does not instantaneously change. Thus, a spiky capacitive current I(C 2 ) occurs, as shown in FIG. 5( e ). The spiky charge (or discharge) current I(C 2 ) of capacitor C 2 , is only limited by the rising (or falling) slope of voltage V(1), the parasitic inductance, the parasitic resistance, and the values of resistors R 1  and R 2 . It is noted that the peak magnitude of the current I(C 2 ) can be very high. If this current flows through the sense resistor Rs, and is not bypassed by bypass diodes D 1  and D 2 , the total voltage on the sense resistor Rs is the combination of the current I(C 2 ) and part of the chopper current I(Q 2 ). In such a situation, the sensing voltage will be significantly distorted, which will affect the control of the chopper operation.  
         [0078]    Actual waveforms, with and without the bypass diodes D 1  and D 2 , are shown in FIGS.  6 ( a ) and  6 ( b ), respectively. As can be seen from the drawings, a positive portion of the current I(C 2 ) flows through I(D 2 ), in which I(C 2 ) is equal to I(D 2 ) when I(C 2 ) is greater than 0. It is noted that a negative portion of the current I(C 2 ) flows through the bypass diode D 1 , which is not shown FIGS.  6 ( a ) and  6 ( b ). A controlling sense voltage, illustrated by the middle trace in FIG. 6( a ), comprises a triangular waveform that exhibits minimal distortion in comparison with the triangular chopper current. When the bypass diodes are omitted, the entire current I(C 2 ) combines with current I(Rs) to form V(Rs), as shown in FIG. 6( b ). A negative portion of the sense voltage almost disappears, due to its combination with the positive portion of the current I(C 2 ). The sensing voltage is significantly distorted in comparison with the chopper current, and the top of the sense voltage is flattened out. The peak of the chopper current that is supposed to be reflected on the sense resistor is not the same as the voltage on the sense resistor. It is noted that it is not possible (or, at least, it is very difficult) to control the peak current of the chopper operation.  
         [0079]    The sensed chopper current, or the current through inductor L 1 , is fed back to control an ON time of chopper switches Q 3  and Q 4 . Further, it is noted that a DC voltage source Vbus is provided by either a power factor correction circuit or directly from a rectified and filtered AC line without power factor correction.  
         [0080]    [0080]FIG. 7( a ) illustrates a portion of the circuit shown in FIG. 1( b ). Resistor R 2  primarily serves as a high impedance path relative to the sense resistor Rs, in order to limit the chopper current flowing through the bypass diodes D 1  and D 2 . A full bridge inverter used with the chopper network forms a triangular current source, as illustrated in FIGS.  7 ( b ) and  7 ( c ). During a normal operation, the operating frequency is much lower than the ignition network, and thus, capacitor C 2  is effectively open (out of the circuit). When the value of resistor R 2  is zero, the maximum voltage on the sense resistor Rs is clamped to approximately 1.4 volts by the conduction of bypass diodes D 1  and D 2 .  
         [0081]    As shown in FIG. 7( b ), the sense voltage V(Rs) exhibits a flat top on its positive waveform, indicating that the waveform is distorted. Bypass diodes D 1  and D 2  will start to conduct, flowing leakage current whenever they are forward biased. The peak sense voltage V(Rs) is generally in the range between approximately 1 Vpk to approximately 2 Vpk. It is noted that it is difficult to accurately control the peak current.  
         [0082]    When the value of resistor R 2  is significantly larger than the value of the sense resistor Rs, current leakage from bypass diodes D 1  and D 2  and resistor R 2  is negligible. The voltage V(Rs) on the sense resistor Rs reflects the true chopper current, as illustrated in FIG. 7( c ). For example, the resistance of the sense resistor Rs is 0.8 ohms for an approximate 70 watt HID lamp with an approximate 1.3 ampere starting current and a peak chopper current of approximately 2 amperes. Based on the formula voltage equals current multiplied by resistance (V=I*R), the peak sense voltage V(Rs) is equal to 2 Amp*0.8 ohms, or 1.6 volts, which is larger than the 1.4 volt clamping voltage of the bypass diodes D 1  and D 2 . Thus, the value of resistor R 2  is chosen to be much larger than the sense resistor (i.e., approximately 20 times the sense resistor Rs, or approximately 15 ohms).  
         [0083]    Resistor R 2  also functions as a damping resistor in series with resistor R 1  when the HID lamp is being started (ignited). Resistor R 2  damps a positive portion of the peak resonant voltage when MOSFET QI is ON when resistor R 2  is connected in the arrangement shown in FIG. 1( b ). In order to dampen both the positive and negative portions of the peak resonant voltage (or just the negative portion of the peak resonant voltage), the connection shown FIG. 1( c ) should be adopted.  
         [0084]    While the invention has been particularly shown and described with reference to the preferred embodiments thereof, it is understood by those skilled in the art that various alterations in form and/or detail may be made without departing from the spirit and/or scope of the invention, as defined by the following claims. For example, an acoustic resonance may be avoided by modifying the present invention to use a high frequency current that is interrupted by a smoothed DC current during a normal operation of a HID lamp. Alternatively, the bridge circuit  10  may be configured as a half bridge circuit.