Abstract:
An apparatus and method for receiving a ranging signal in an OFDMA mobile communication system are provided. The ranging signal receiving apparatus including, a ranging subchannel extractor for extracting subcarrier values with a ranging signal from a (FFT) signal; a plurality of multipliers for code-demodulating the sub-carrier values by multiplying them by a plurality of ranging codes; each of a plurality of correlators for calculating a plurality of differential correlations in a code-demodulated signal received from a corresponding multiplier; each of a plurality of inverse fast Fourier transform (IFFT) processors for IFFT-processing differential correlations received from a corresponding correlator by mapping the differential correlations to predetermined subcarriers and each of a plurality of maximum value detectors for detecting a maximum value in an IFFT signal received from a corresponding IFFT processor and calculating a timing offset using an IFFT output index having the maximum value.

Description:
PRIORITY  
       [0001]     This application claims priority under 35 U.S.C. § 119 to an application entitled “Apparatus And Method For Detecting Ranging Signal In An Orthogonal Frequency Division Multiple Access Mobile Communication System” filed in the Korean Intellectual Property Office on Oct. 12, 2004 and assigned Ser. No. 2004-81326, the contents of which are incorporated herein by reference.  
       BACKGROUND OF THE INVENTION  
       [0002]     1. Field of the Invention  
         [0003]     The present invention relates generally to a receiving apparatus and method for a base station (BS) in an Orthogonal Frequency Division Multiplexing (OFDM)-based broadband mobile communication system, and more particularly, to an apparatus and method for receiving a ranging signal in an Orthogonal Frequency Division Multiple Access (OFDMA) communication system.  
         [0004]     2. Description of the Related Art  
         [0005]     In a communication system which is defined by an Institute of Electronics and Electrical Engineers (IEEE) 802.16d/e standard, a BS acquires uplink timing synchronization and tracks Carrier-to-Interference plus Noise Ratio (CINR) using a known signal (e.g. a ranging signal, a preamble, a pilot signal, etc.) received from a subscriber station (SS). A signal that the SS transmits to help the BS to acquire the uplink timing synchronization is known as a “ranging signal”. Conventional ranging signal reception will now be described, according to the IEEE 802.16d/e standard.  
         [0006]      FIG. 1  is a block diagram schematically illustrating the configuration of an OFDMA-based broadband mobile communication system. The OFDMA communication system is configured to have a single cell structure, and includes a BS  100  and a plurality of SSs  110 ,  120  and  130  managed by the BS  100 . Signal transmission/reception takes place using an OFDM/OFDMA based communication scheme between the BS  100  and the SSs  110 ,  120  and  130 . Thus, the SSs  110 ,  120  and  130  and the BS  100  transmit physical channel signals on subcarriers.  
         [0007]     OFDMA defines an access scheme of a two-dimensional grid that combines Time Division Access (TDM) with Frequency Division Access (FDM). In OFDMA, data symbols are delivered on subcarriers which form subchannels. Depending on system situation, a predetermined number of subcarriers form one subchannel.  
         [0008]     For application of Time Division Duplexing (TDD) to the OFDMA communication system, ranging is required to acquire accurate timing synchronization between the SS and the BS and adjust the reception power of the BS on the uplink. In each OFDMA frame a ranging channel has a plurality of subchannels for transmitting a ranging signal.  
         [0009]     Ranging in the IEEE 802.16d/e communication system will be described below. The ranging is classified into initial ranging for acquiring physical layer timing synchronization and periodic ranging for maintenance and management.  
         [0010]     The initial ranging is the process of acquiring a correct timing offset between the BS and the SS and initially adjusting a transmit power. Upon power-on, the SS acquires downlink synchronization from a received downlink preamble signal. Then the SS performs the initial ranging with the BS to adjust an uplink time offset and transmit power. The IEEE 802.16d/e communication systems use the OFDM/OFDMA communication scheme. Thus, they perform a ranging procedure by transmitting a randomly selected ranging code on a plurality of subchannels.  
         [0011]     The periodic ranging is the process of periodically tracking the uplink timing offset and received signal strength after the initial ranging. The SS randomly selects one of ranging codes allocated for the periodic ranging in the ranging procedure.  
         [0012]     A description of transmitting a ranging signal will now be provided.  
         [0013]      FIG. 2  is a block diagram illustrating a ranging code generator used in a typical TDD/OFDMA system. A Pseudorandom Noise (PN) code generated from a Pseudo Random Binary Sequence (PRBS) generator is used as a ranging code. The generator polynomial for generating a PN code is given as 
   G ( x )=1+ x   1   +x   4   +x   7   +x   15   Equation 1 
         [0014]     A register is initialized to 00101011 (binary) and a 7-bit cell identification (ID) number. The SS acquires the cell ID number from a downlink preamble signal or broadcast information.  
         [0015]     For a ranging code length of N bits, codes are generated for each ranging mode as follows.  
         [0016]     A long sequence is generated under synchronization of 1360 th  through (N×K1) th  clock pulses from the PRBS generator. The long sequence is divided into K1 N-bit codes for use in initial ranging. For handoff ranging, a long sequence generated under synchronization of (N×K1+1) th  through N×(K1+K2) th  clock pulses from the PRBS generator is divided into K2 N-bit codes. K3 N-bit codes are used for periodic ranging, which are created by dividing a long sequence generated under synchronization of N×(K1+K2+1) th  through N×(K1+K2+K3) th  clock pulses from the PRBS generator by N bits. For bandwidth request ranging, a long sequence generated under synchronization of (N×K1+K2+K3+1) th  through N×(K1+K2+K3+K4) th  clock pulses from the PRBS generator is divided into K4 N-bit codes. (K1, K2, K3 and K4 are number of codes).  
         [0017]      FIG. 3  is a block diagram illustrating a ranging transmitter in an SS in a conventional TDD/OFDMA communication system.  
         [0018]     Referring to  FIG. 3 , upon receipt of information about an SS-intended ranging mode (e.g. initial ranging, periodic ranging, etc.), a ranging code generator  301  generates a randomly selected ranging code. A ranging channel generator  302  allocates the ranging code to subcarriers. The subcarrier allocation amounts to providing each element or bit of the ranging code to a corresponding input (subcarrier position) of anInverse Fast Fourier Transform (IFFT) processor  303 . 0s are padded at subcarrier positions to which the ranging code is not allocated. The IFFT processor  303  generates time-domain signals by IFFT-processing the signal from the ranging channel generator  302 . A parallel-to-serial (P/S) converter  304  converts the parallel time-domain signals to serial data. A Cyclic Prefix (CP) inserter  305  inserts a CP into the data stream, thereby creating a baseband ranging signal. While not shown, the baseband ranging signal is processed into a transmittable Radio Frequency (RF) signal and wirelessly transmitted through an antenna.  
         [0019]     A ranging channel pattern as defined by the IEEE 802.16e is illustrated in  FIG. 4  in which a total of 144 tones (subcarriers) used for transmission of the ranging signal reside in six bands that are separated from each other, each band including 24 successive subcarriers.  
         [0020]     Reception of the ranging signal will be described below.  
         [0021]      FIG. 5  is a block diagram illustrating a ranging receiver in a BS in the conventional TDD/OFDMA communication system.  
         [0022]     Referring to  FIG. 5 , a Fast Fourier Transform (FFT) processor  501  FFT-processes an input signal and outputs the resulting frequency-domain signal. That is, the FFT processor  501  demodulates the input signal to subcarrier values. A ranging subchannel extractor  502  extracts subcarrier values with a ranging code loaded thereon from the subcarrier values received from the FFT processor  501 . A multiplier  503  multiplies the extracted subcarrier values by ranging code 0 (or Code 0). A multiplier  504  multiplies the extracted subcarrier values by ranging code 1 (Code 1). Similarly, a multiplier  505  multiplies the extracted subcarrier values by ranging code (k−1) (Code (k−1)). Without knowledge of a received ranging code, all possible ranging codes are multiplied by the subcarrier values with the ranging code.  
         [0023]     A phase detector  506  detects a timing offset from the product received from the multiplier  503 . A phase detector  507  detects a timing offset from the product received from the multiplier  504 . Similarly, a phase detector  508  detects a timing offset from the product received from the multiplier  505 . The operations of the phase detectors  506  to  508  are modeled as defined by Equation 2 below.  
               ℜ   ⁡     (   n   )       =       argmax         t   min     /     θ   step       ≤   n   ≤       t   max     /     θ   step           ⁢       ∑             m   ∈     {     0   ,   M     }       ,     RNG   subband                   k   ∈     {     0   ,     K   -   1       }       ,       tone   ⁢           ⁢   index     -   in   -   subband               ⁢       Y     m   ,   k       ⁢     C     m   ,   k       ⁢     ⅇ       -   j     ⁢           ⁢   2   ⁢   π   ⁢           ⁢     f   ⁡     (     m   ,   k     )       ×       (     n   ⁢           ⁢     θ   step       )     /     N   FFT                         Equation   ⁢           ⁢   2             
 
 where Y m,k  denotes the received signal response of a k th  subcarrier in an m th  band in  FIG. 4 , C m,k  denotes a ranging code bit allocated to the k th  subcarrier in the m th  band, f(m,k) denotes the frequency index of the k th  subcarrier in the m th  band, N FFT  denotes an FFT size (for example 1024), and θ step  denotes samples normalized to a step size (expressed in the number of samples normalized to a sampling rate) set for timing offset detection. 
 
         [0024]     In Equation. 2, {Y m,k , C m,k ,} is the product of the FFT processor output by a ranging code, input to a phase detector. This value is multiplied by an exponential function. A variable set in the exponential function is n and n ranges [t min /θ step □ t max /θ step ]. n denotes a timing offset range to be estimated. Using Equation 2, {         (n), t min /θ step ≦n≦t max /θ step } is computed over all possible values of n. An n value that maximizes |R(n)| is selected as a temporary timing offset, n est .  
         [0025]     Peak detectors  509  to  511  each calculate a Peak-to-Average Power Ratio (PAPR) to verify the temporary timing offset received from a corresponding phase detector and compare the PAPR with a predetermined threshold. If the PAPR is greater than the threshold, the temporary timing offset is decided as a timing offset estimate. If the PAPR is less than the threshold, the temporary timing offset is discarded and it is determined that a ranging signal has not been received.  
         [0026]     The PAPR is computed using Equation 3 below.  
             PAPR   =              ℜ   ⁡     (     n   est     )            2       average   ⁢     {              ℜ   ⁡     (   n   )            2     ,         t   min     /     θ   step       ≤   n   ≤       t   max     /     θ   step           }                 Equation   ⁢           ⁢   3             
 
         [0027]     As described above, the conventional TDD/OFDMA communication system detects a ranging signal in the manner illustrated in  FIG. 5 , and suffers from the following problems.  
         [0028]     (1) Acutal implementation is difficult because of computational complexity.  
         [0029]     The FFT processor  501  and the multipliers  503  to  505  are basic computation blocks and the phase detectors  506  to  508  detect phases using Equation 2. As noted from Equation 2, 1024 exponential calculations are performed on a value received from a multiplier for one n value and accumulated. Then a maximum value is selected as a temporary timing offset. The peak detectors  509  to  511  calculate PAPRs to verify the temporary timing offsets. The implementation complexity is illustrated in Table 1 below.  
                                   TABLE 1                           FFT                           reception       Real   (Radi × 2   Code           Total       multiplication   FFT)   Multiplication   Phase Test   Peak Test   computation                   Conventional   N FFT log 2 N FFT     2 × Number_of_Codes ×   2 × Number_of_Codes ×   2 × Number_of_Codes ×   9.46E6               Code_Size   Code_Size × N FFT     Code_Size                 In Table 3 it is assumed that:            N FFT : FFT size (e.g., 1024)            Number_of_Codes: the number of ranging codes (e.g., 32)            Code_Size: the length of ranging codes (e.g., 144).             
 
         [0030]     As illustrated in Table 1, according to the IEEE 802.16e, 3 (ranging type)×9.46E6 (computation volume)=28.4E6 real multiplications occur every 5 msec, or 5679E6 floating point calculations take place every second. Therefore, the conventional ranging detection is very difficult to implement.  
         [0031]     (2) Ranging reception performance decreases at low Carrier-to-Interference plus Noise Ratio (CINR). Since the ranging channel is not transmitted over the total frequency band, the timing offset estimation can be incorrect.  
         [0032]     To be more specific, conventionally, the response of a channel whose phase is rotated by a timing offset in the frequency domain is achieved and then converted to a time-domain channel response, thereby detecting the shift of the time-domain channel response. As described earlier with reference to  FIG. 4 , since the ranging code is loaded only in some bands, the frequency characteristic of an acquired channel is limited. Meanwhile, conversion of a channel value to the time domain is equivalent to passing through a filter configured in correspondence with a ranging subchannel. Therefore, the output of the phase detector is the convolution of the time response of an ideal channel with a filter coefficient. That is, the phase detector outputs an incorrect timing offset. Considering the effects of noise, the performance is worsened. In a cellular system, many terminals must operate at a low CINR due to inter-cell interference. Since the CINR is a function of distance in constant transmit power and the same path loss, abnormal ranging reception at a low CINR reduces cell radius.  
       SUMMARY OF THE INVENTION  
       [0033]     An object of the present invention is to substantially solve at least the above problems and/or disadvantages and to provide at least the advantages below. Accordingly, an object of the present invention is to provide an apparatus and method for reducing a computation requirement for ranging signal detection in an OFDMA mobile communication system.  
         [0034]     Another object of the present invention is to provide an apparatus and method for improving the performance of detecting a ranging signal in an OFDMA mobile communication system.  
         [0035]     The above objects are achieved by providing an apparatus and method for receiving a ranging signal in an OFDMA mobile communication system.  
         [0036]     According to an embodiment of the present invention, in a base station (BS) apparatus of a broadband mobile communication system, a ranging subchannel extractor extracts subcarrier values with a ranging signal from an FFT signal. A plurality of multipliers code-demodulate the sub-carrier values by multiplying them by a plurality of ranging codes. Each of a plurality of correlators calculates a plurality of differential correlations in a code-demodulated signal received from a corresponding multiplier. Each of a plurality of IFFT processors IFFT-processes differential correlations received from a corresponding correlator by mapping the differential correlations to predetermined subcarriers. Each of a plurality of maximum value detectors detects a maximum value in an IFFT signal received from a corresponding IFFT processor and calculates a timing offset using an IFFT output index having the maximum value.  
         [0037]     According to another aspect of the present invention, in a receiving method in a base station of a broadband mobile communication system, subcarrier values with a ranging signal are extracted from an FFT signal. The sub-carrier values are multiplied by a plurality of ranging codes, for code modulation. A plurality of differential correlations are calculated for each of the code-demodulated signals and IFFT-processed by mapping the differential correlations to predetermined subcarriers. A maximum value is detected in each of the IFFT signals and a timing offset is calculated using an IFFT output index having the maximum value. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0038]     The above and other objects, features and advantages of the present invention will become more apparent from the following detailed description when taken in conjunction with the accompanying drawings in which:  
         [0039]      FIG. 1  schematically illustrates the configuration of an OFDMA-based broadband mobile communication system;  
         [0040]      FIG. 2  illustrates a ranging code generator in a typical TDD/OFDMA communication system;  
         [0041]      FIG. 3  is a block diagram illustrating a ranging transmitter in an SS in a conventional TDD/OFDMA communication system;  
         [0042]      FIG. 4  illustrates a ranging channel pattern in the typical TDD/OFDMA communication system;  
         [0043]      FIG. 5  is a block diagram illustrating a ranging receiver in a BS in the conventional TDD/OFDMA communication system;  
         [0044]      FIG. 6  is a block diagram illustrating a ranging receiver in a BS in a TDD/OFDMA communication system according to an embodiment of the present invention;  
         [0045]      FIG. 7  illustrates a J-point IFFT processor and its inputs according to an embodiment of the present invention; and  
         [0046]      FIG. 8  is a flowchart illustrating a ranging signal detection operation in the BS in the TDD/OFDMA communication system according to the embodiment of the present invention. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT  
       [0047]     A preferred embodiment of the present invention will be described herein below with reference to the accompanying drawings. In the following description, well-known functions or constructions are not described in detail since they would obscure the invention in unnecessary detail.  
         [0048]     The present invention is intended to provide a method of reducing a computation requirement for ranging signal detection and improving ranging detection performance even at a low CINR in an OFDMA mobile communication system. In the OFDMA mobile communication system, an SS transmits a predetermined signal such as a ranging signal, a pilot signal or a preamble signal to a BS, for uplink synchronization.  
         [0049]     The present invention as described below is applicable without limitation to any TDD-OFDMA system that acquires an uplink synchronization using a predetermined signal such as a ranging signal.  
         [0050]      FIG. 6  is a block diagram illustrating a ranging receiver in a BS in a TDD/OFDMA communication system according to an embodiment of the present invention.  
         [0051]     Referring to  FIG. 6 , an FFT processor  601  FFT-processes a received signal and outputs the resulting frequency-domain signal. That is, the FFT processor  601  demodulates the received signal to subcarrier values. A ranging subchannel extractor  602  extracts subcarrier values with a ranging code among the subcarrier values. A multiplier  603  multiplies the extracted subcarrier values by ranging code 0 (or Code 0). A multiplier  604  multiplies the extracted subcarrier values by ranging code 1 (Code 1). Similarly, a multiplier  605  multiplies the extracted subcarrier values by ranging code (k−1) (Code (k−1)). In this way, the subcarrier values with the ranging code are multiplied by all possible ranging codes (i.e., K codes).  
         [0052]     The output Y m,k C m,k  of the multipliers  603  to  605  represents the frequency characteristic of a channel that the ranging signal has experienced in the case in which physical ranging signals have not collided, and contains a phase rotation component arising from a generated timing offset. Y m,k  denotes the received signal response of a k th  subcarrier in an m th  band and C m,k  denotes a ranging code bit allocated to the k th  subcarrier in the m th  band as shown in  FIG. 4 .  
         [0053]     A correlator (or differential correlator)  606  groups values received from the multiplier  603  according to ranging bands, calculates differential correlations between two subcarriers spaced apart from each other by k (1≦k ≦k   max ) (k is a IFFT input index) over all cases in each ranging band, and sums the differential correlations for each k value across the ranging bands, thereby creating k th -order differential correlations. Then the correlator  606  finally produces 2×k max  correlations by complex-conjugating the k th -order differential correlations. Each correlation Z k  output from the correlator  606  is the sum of differential correlations between subcarriers spaced apart from each other by k, including a phase rotation component corresponding to an uplink timing offset.  
         [0054]     In the same manner, the correlator  608  groups values received from the multiplier  605  according to the ranging bands, calculates differential correlations between two subcarriers apart from each other by k (1≦k≦k max ) over all cases in each ranging band, and sums the differential correlations for each k value across the ranging bands, thereby creating k th -order differential correlations. Then the correlator  608  finally produces 2×k max  correlations by complex-conjugating the k th -order differential correlations.  
         [0055]     The operation of the correlators  606  to  608  are each defined by Equation 4 below.  
               Z   k     =     {                   ∑     l   =   0     5     ⁢       ∑     n   =   0       23   -   k       ⁢       (       Y     l   ,   n       ⁢     C     l   ,   n         )     ⁢       (       Y     l   ,     m   +   k         ⁢     C     l   ,     n   +   k           )     *           ,           l   ≤   k   ≤     k   max                   Z     J   -   k     *     ,             J   -     k   max       ≤   k   &lt;   J           ⁢   Z   ⁢     
     ⁢   where     ,     
     ⁢       Z   k     ⁢     :     ⁢   J   ⁢     -     ⁢   point   ⁢           ⁢   IFFT   ⁢           ⁢   complex   ⁢           ⁢   input   ⁢           ⁢   value   ⁢     
     ⁢   k   ⁢     :     ⁢   J   ⁢     -     ⁢   point   ⁢           ⁢   IFFT   ⁢           ⁢   input   ⁢           ⁢   index     ,     0   ≤   k   &lt;     J   ⁢     -     ⁢   point                   Equation   ⁢           ⁢   4             
 
         [0056]     Equation 4 is based on the assumption that values corresponding to six ranging bands each having 24 subcarriers, that is, 144 frequency-domain values are fed to each correlator. Z k  is defined as the sum of correlations between subcarriers separated from each other by k. If the subcarriers spaced by k have the same channel characteristics, the amplitude of Z k  is the sum of channel amplitudes, and its phase is the difference between the phases of subcarriers apart from each other by k affected by a timing offset. The number of summing (Σ) operations varies depending on a k value. This is related to the reliability of information. As k decreases, the correlation between adjacent subcarriers is higher. Accordingly, as the number of summing operations increase, the value of Z k  also increases in as defined by Equation 4. Therefore, the reliability of Z k  is increased. Each ranging band includes 24 successive subcarriers, 23 Z k  values are available since k ranges from 1 to 23. Although a phase difference can be obtained with a negative value of k, the phase difference is equivalent to the complex conjugate of Z k . Hence, Z k  for k ranging from −1 to −23 is easily achieved without re-computing Equation 4. As a result, a total of 46 Z k  values are output from each correlator. These Z k  values are symmetrical in the form of a triangle centering on 0.  
         [0057]     Each of zero padders  609  to  611  provides the 2×k max  correlations received from a corresponding correlator to appropriate inputs of a corresponding J-point IFFT processor and pads zeros in non-allocated inputs of the IFFT processor. For k max =23, zero-padding positions Z k  are defined by Equation 5. 
 
 Z   k =0 , k= 0, 24 ≦k&lt;j− 24  Equation 5 
 
         [0058]     J-point IFFT processors  612  to  614  IFFT-process signals received from their corresponding zero padders  609  to  611  and output time-domain signals. In the present invention, the IFFT size J can be selected from 
 
Jε{2 3 ,2 4 ,2 5 , . . . , N FFT }
 
         [0059]      FIG. 7  illustrates a J-point IFFT processor and its inputs according to an embodiment of the present invention.  
         [0060]     Referring to  FIG. 7 , the inputs of the J-point IFFT processor are {Z 0 , Z 1 , . . . , Z J/2−1 , Z J/2 , Z J/2+1 , . . . , Z J−2 , Z J−1 }. The output of the J-point IFFT processor is the square of a sin c function due to the waveform of the input signal Z k , characteristic of a shifted maximum value caused by the uplink timing offset.  
         [0061]     Therefore, maximum value detectors  615  to  617  (as shown in  FIG. 6 ) each detects a maximum value from the signal |sin c| 2  received from a corresponding J-point IFFT processor and calculates a temporary timing offset using an IFFT output index with the maximum value.  
         [0062]     Let the output of the J-point IFFT processor be denoted by z n . Then, the maximum value detector operates as defined by Equation 6 below.  
               n   =       argmax     0   ≤   n   ≤     J   -   1         ⁢     {            z   n          2     }         ⁢     
     ⁢       Δ   ⁢           ⁢     t   offset       =     {                 DR   ×   n     ,             if   ⁢           ⁢   n     ≤     j   2                     DR   ×   n     -     N   FFT       ,             if   ⁢           ⁢   n     &gt;     j   2             ⁢     
     ⁢   where   ⁢     
     ⁢     DR   ⁡     (     decimation   ⁢           ⁢   ratio     )         =       N   FFT     J                   Equation   ⁢           ⁢   6             
 
         [0063]     Each of PAPR comparators  618  to  620  calculates a PAPR using Equation 7 to verify the temporary timing offset received from a corresponding maximum value detector, and compares the PAPR with a predetermined threshold. If the PAPR exceeds the threshold, the PAPR comparator outputs the temporary timing offset as a timing offset estimate Δt offset,final .  
               Δ   ⁢           ⁢     t     offset   ,   final         =     {               Δ     offset   ,               if   ⁢           ⁢   PAPR     ≥   threshold                 N   /   A     ,         others         ⁢           ⁢   where   ⁢     
     ⁢   threshold   ⁢     :     ⁢           ⁢   the   ⁢           ⁢   specific   ⁢           ⁢   value   ⁢           ⁢   assigned   ⁢           ⁢   to   ⁢           ⁢   BS   ⁢     
     ⁢   PAPR     =       max   ⁢     {            IFFT   ⁢     {     Z   k     }            2     }         average   ⁢     {            IFFT   ⁢     {     Z   k     }            2     }                     Equation   ⁢           ⁢   7             
 
         [0064]      FIG. 8  is a flowchart illustrating a ranging detection operation in the BS in the TDD/OFDMA communication system according to the embodiment of the present invention.  
         [0065]     Referring to  FIG. 8 , the BS demodulates a received signal to subcarrier values using an FFT in step  801  and multiplies the subcarriers by all possible ranging codes in step  803 .  
         [0066]     In step  805 , the BS groups each of the ranging code-demodulated signals according to ranging bands, calculates differential correlations between subcarriers spaced apart from each other by k (1≦k≦k max ) over all possible cases in each ranging band, and sums the differential correlations for each k value across the ranging bands, resulting in k th -order differential correlations, and then complex-conjugates the k th -order differential correlations. Thus, 2×k max  correlations are produced for each ranging code-demodulated signal. For 6 ranging bands each having 24 subcarriers, let the received signal response of an n th  subcarrier in an 1 th  band be denoted by Y 1,n  and the ranging code bit allocated to the n th  subcarrier in the 1 th  band be denoted by C 1,n . Then 2×k max  correlations calculated for one ranging code-demodulated signal are computed using Equation 8 below.  
               Z   k     =     {                   ∑     l   =   0     5     ⁢       ∑     n   =   0       23   -   k       ⁢       (       Y     l   ,   n       ⁢     C     l   ,   n         )     ⁢       (       Y     l   ,     m   +   k         ⁢     C     l   ,     n   +   k           )     *           ,           l   ≤   k   ≤     k   max                   Z     J   -   k     *     ,             J   -     k   max       ≤   k   &lt;   J           ⁢     
     ⁢   where     ,     
     ⁢       Z   k     ⁢     :     ⁢   J   ⁢     -     ⁢   point   ⁢           ⁢   IFFT   ⁢           ⁢   complex   ⁢           ⁢   input   ⁢           ⁢   value   ⁢     
     ⁢   k   ⁢     :     ⁢   J   ⁢     -     ⁢   point   ⁢           ⁢   IFFT   ⁢           ⁢   input   ⁢           ⁢   index     ,     0   ≤   k   &lt;     J   ⁢     -     ⁢   point                   Equation   ⁢           ⁢   8             
 
 where k max  is 23 because each band has 24 successive subcarriers. 
 
         [0067]     In step  807 , the BS allocates the 2×k max  correlations for each ranging code to subcarriers. At the same time, subcarriers without the correlations are padded with zeroes. For example, if k max =23, zero-padded subcarriers Z k  are determined using Equation 9 below. 
 
 Z   k =0 , k= 0, 24 ≦k&lt;j− 24  Equation 9 
 
         [0068]     After the subcarrier allocation, the BS performs a J-point IFFT operation on each of the subcarrier-allocated signals in step  809 . The IFFT size J is a system operation parameter. The resulting IFFT signal is the square of a sin c function has a shifted maximum value according to a timing offset.  
         [0069]     Therefore, the BS detects a maximum value from each IFFT signal and calculates a timing offset using an IFFT output index with the maximum value in step  811 . 
 
 If the IFFT signal is z n , the timing offset is computed using Equation 10 below.  
               n   =       argmax     0   ≤   n   ≤     J   -   1         ⁢     {            z   n          2     }         ⁢     
     ⁢       Δ   ⁢           ⁢     t   offset       =     {                 DR   ×   n     ,             if   ⁢           ⁢   n     ≤     j   2                     DR   ×   n     -     N   FFT       ,             if   ⁢           ⁢   n     &gt;     j   2             ⁢     
     ⁢   where   ⁢     
     ⁢     DR   ⁡     (     decimation   ⁢           ⁢   ratio     )         =       N   FFT     J                   Equation   ⁢           ⁢   10             
 
         [0070]     In step  813 , the BS calculates the PAPR of each IFFT signal using Equation 11 below.  
               Δ   ⁢           ⁢     t     offset   ,   final         =     {               Δ     offset   ,               if   ⁢           ⁢   PAPR     ≥   threshold                 N   /   A     ,         others         ⁢     
     ⁢   where   ⁢     
     ⁢   threshold   ⁢     :     ⁢           ⁢   the   ⁢           ⁢   specific   ⁢           ⁢   value   ⁢           ⁢   assigned   ⁢           ⁢   to   ⁢           ⁢   BS   ⁢     
     ⁢   PAPR     =       max   ⁢     {            IFFT   ⁢     {     Z   k     }            2     }         average   ⁢     {            IFFT   ⁢     {     Z   k     }            2     }                     Equation   ⁢           ⁢   11             
 
         [0071]     The BS then compares the PAPR with a predetermined threshold in step  815 . If the PAPR exceeds the threshold, the BS decides a timing offset corresponding to the PAPR as a timing offset estimate Δt offset,final  and stores the timing offset and its associated ranging code in step  817 . If the PAPR is less than the threshold, the BS discards the timing offset.  
         [0072]     Compared to the conventional ranging detection method, the ranging method according to present invention provides better reception performance. A comparison in reception performance between the conventional technology and the present invention is given in Table 2 below.  
                                                                                     TABLE 2                                                   Veh A,   Veh B,           AWGN   Ped A, 3 Km/h   Ped B, 10 Km/h   60 Km/h   120 Km/h            CINR   Conventional   Present   Conventional   present   Conventional   present   Conventional   present   Conventional   present               −5 dB     1.0000   0.9989   0.9995   0.9999   0.6578   0.9304   0.8732   0.9259   0.7959   0.8480       0 dB   1.0000   1.0000   1.0000   1.0000   0.9171   0.9996   0.9731   0.9995   0.9557   0.9609       5 dB   1.0000   1.0000   1.0000   1.0000   0.9306   1.0000   0.9789   1.0000   0.9572   0.9724                  
 
         [0073]     CINR denotes a Carrier-to-Interference plus Noise Ratio, AWGN denotes Additive White Gaussian Noise, PED Denotes a pedestrian environment and Veh denotes a Vehicular environment. Table 3 below illustrates reception ranging reception performance for each J-point IFFT size according to the present invention.  
                                                                     TABLE 3                           IFFT       Ped A,   Ped B,   Veh A,   Veh B,       CINR   size   AWGN   3 Km/h   10 Km/h   60 Km/h   120 Km/h                                −5 dB     64   0.9949   0.9984   0.8972   0.8931   0.8016           128   0.9980   0.9992   0.9247   0.9178   0.8390           256   0.9989   0.9999   0.9304   0.9259   0.8480           512   0.9987   0.9999   0.9294   0.9250   0.8492       0 dB   64   1.0000   1.0000   0.9988   0.9994   0.9441           128   1.0000   1.0000   0.9995   0.9997   0.9579           256   1.0000   1.0000   0.9996   0.9995   0.9609           512   1.0000   1.0000   0.9992   0.9996   0.9596       5 dB   64   1.0000   1.0000   0.9998   1.0000   0.9559           128   1.0000   1.0000   1.0000   1.0000   0.9712           256   1.0000   1.0000   1.0000   1.0000   0.9724           512   1.0000   1.0000   1.0000   1.0000   0.9738                  
 
         [0074]     Particularly, the present invention is less complex and requires fewer computations than the conventional technology, as illustrated in Table 4 below.  
                                       TABLE 4                           FFT                               reception               Total   Total       Real   (Radi × 2   Code       IFFT   computation   computation       multiplication   FFT)   Multiplication   Diff. demod   (Radi × 2)   N J  = 126   N J  = 256                   Present   N FFT log 2 N FFT     2 × Num_of_Codes ×   Num_of_Codes × 3312   Num_of_Codes ×   1.09E6   2.07E6       invention       Code_Size       N J log 2 N J                   Where it is assumed that:            N FFT : FFT size (e.g., 1024)            Number_of_Codes: the number of ranging codes (e.g., 32)            Code_Size: the length of ranging codes (e.g., 144).             
 
         [0075]     As illustrated in Table 4, for an N j -IFFT size of 126, the computation volume is 1.09E6 and for an N j -IFFT size of 256, the computation volume is 2.07E6 in the present invention. On the other hand, the conventional technology has a computation volume of 9.46E6 as illustrated in Table 1, which is about 900% of the computation volume of the present invention.  
         [0076]     As described above, the present invention advantageously improves the reception performance of a ranging signal and reduces a computation requirement for ranging signal detection.  
         [0077]     While the invention has been shown and described with reference to a certain preferred embodiment thereof, it will be understood by those skilled in the art that various changes in form and details may be made therein without departing from the spirit and scope of the invention as defined by the appended claims.