Abstract:
One or more waveform generators impose an input waveform across touch sensor components. Phase shift detection logic measures a phase shift between the input waveform and an output waveform component associated with each waveform generator appearing across a touch sensor component. Amplitude ratio detection logic measures an output-to-input waveform magnitude ratio associated with each waveform generator. Translator logic converts the phase shift(s) and optionally the output-to-input waveform magnitude ratio(s) to capacitance values. Validation logic compares the resulting capacitance values to each other to determine consistency and validity during a particular waveform sampling time. The phase shift detection logic and the amplitude ratio detection logic are optionally implemented with discrete Fourier transform logic which includes narrow-band filtering to exclude electromagnetic interference frequency components from touch sensor signal measurements.

Description:
PRIORITY CLAIM 
       [0001]    This application claims the benefit of priority under 35 U.S.C. §119(e) to U.S. Provisional Patent Application Ser. No. 61/856,335 titled “Robust Capacitive Sensing,” filed on Jul. 19, 2013 and to U.S. Provisional Patent Application Ser. No. 61/877,760 titled “Robust Capacitive Sensing” filed on Sep. 1, 2013, both incorporated herein by reference in their entirety. 
     
    
     TECHNICAL FIELD 
       [0002]    Structures and methods described herein relate to capacitive touch sensing, including the generation, detection, and processing of capacitive sensor signals to achieve higher signal-to-noise ratios, greater sensitivity, and increased reliability of capacitance measurements in the presence of a background electromagnetic interference field. 
       BACKGROUND INFORMATION 
       [0003]    Capacitive sensors utilize a conductive element and a proximate ground plane or one or more proximate conductive elements separated by a dielectric material to form one or more capacitive touch elements. Each capacitive touch element exhibits a capacitance that can vary according to a proximity of a mass referred to herein as a “dielectric mass.” The dielectric mass may be any material proximate to the capacitive touch element which changes the overall dielectric strength of the capacitive touch element. For example, a human finger or other animal digit or body part, a fluid flow, an article of manufacture passing by on an assembly line, a tooth of a rotating gear and the like may all constitute dielectric masses. It is noted that the terms “capacitive sensor” and “capacitive touch sensor” are used interchangeably herein. For purposes of this disclosure, the latter term is not to be construed as requiring physical contact of a human digit with the sensor. 
         [0004]    A capacitive sensor may consist of a single capacitive touch element or multiple capacitive touch elements. A single touch element sensor may be used in applications ranging from a simple on/off switch to counting applications and relative measurement of a distance of one or more proximate dielectric masses to the touch element. Touch elements of a multi-element capacitive sensor may be organized into a two-dimensional matrix and used to sense a position of a dielectric mass relative to the multi-element matrix. 
         [0005]    Electronic circuitry used to detect, amplify, filter and otherwise process capacitive sensor signals is referred to herein as “capacitive touch detection apparatus.” Capacitive touch detection apparatus generates one or more output signals or values as relative or absolute measures of capacitance indicating the proximity of a dielectric mass to the sensor. Capacitive sensors and capacitive touch detection apparatus are traditionally susceptible to electromagnetic interference (“EMI”) due to their high impedance characteristics. Such EMI may include power line noise, energy-saving lamp noise, noise from switching power supplies, clock and data line noise, and so forth. Fundamental and harmonic frequencies associated with EMI may overlap frequencies of capacitive sensor signals. Such overlap may cause the capacitive touch detection apparatus to output erroneous values of capacitance and to consequently mis-interpret the proximity or position of a dielectric mass to the sensor. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0006]      FIG. 1  is a schematic diagram of a capacitive touch detection apparatus according to various example embodiments of the invention. 
           [0007]      FIG. 2  is a waveform diagram of input and output voltage waveforms associated with a capacitive touch detection apparatus according to various example embodiments. 
           [0008]      FIG. 3  is a schematic diagram of a capacitive touch detection apparatus according to various example embodiments. 
           [0009]      FIG. 4  is a schematic diagram of a capacitive touch detection apparatus according to various example embodiments. 
           [0010]      FIG. 5  is a frequency plot illustrating a filtering effect associated with a capacitive touch detection apparatus according to various example embodiments. 
           [0011]      FIGS. 6A-6C  are a flow diagram illustrating an example method of capacitive touch detection according to various example activities. 
       
    
    
     SUMMARY OF THE INVENTION 
       [0012]    Apparatus and methods disclosed herein generate one or more periodic input waveforms of frequencies F1, F2 . . . FN and impose the input waveforms across a series-connected resistive-capacitive (“RC”) network. The capacitor in the RC network is a capacitive touch element. The RC network causes both a voltage drop and a phase shift of the periodic waveform(s) as measured across either the capacitive touch element or the resistor as output waveform(s). A phase shift detector measures the phase shift between the input and output waveforms. A phase shift to capacitance translator determines the instantaneous capacitance of the capacitive touch element as a function of the phase shift. 
         [0013]    In some embodiments, an amplitude ratio detector measures the ratio of the amplitude of the output waveform to the amplitude of the input waveform. An amplitude ratio to capacitance translator determines an additional measure of the instantaneous capacitance of the capacitive touch element as a function of the ratio of the amplitude of the output waveform to the amplitude of the input waveform. In some embodiments, validation logic outputs one or more of the measures of the instantaneous capacitance or generates an average or weighted average of the one or more measures if the difference between the measures does not exceed a specified value. In some embodiments, the validation logic outputs one or more of the measures of the instantaneous capacitance or an average or weighted average of the one or more measures without regard to the difference between the one or more measures. 
         [0014]    In some embodiments, the output waveform phase and amplitude are measured by performing analog-to-digital (“ADC”) conversions of the output waveform. A sampling clock used to clock the ADC may also be used to synchronize the waveform generator(s) used to generate the periodic input waveforms. A set of output waveform magnitude values sampled from portions of a cycle of a periodic input waveform is acted upon by a discrete Fourier transform (“DFT”) logic module associated with a waveform generator of a particular frequency. In some embodiments the DFT is a single-bin DFT (“SB DFT”). DFT embodiments have the advantage of creating a narrow-band filter around the waveform generator frequency and thus excluding previously-discussed EMI frequency components outside of the narrow-band filter. Embodiments including two or more periodic waveform generators, each of a different frequency, may use capacitance values determined from measurements utilizing a waveform generator frequency not subject to unacceptable levels of EMI frequency components at the time of a particular instantaneous touch element capacitance measurement. 
         [0015]    Real (“I”) and imaginary (“Q”) outputs of each DFT logic module are operated on by phase shift logic to determine the instantaneous phase of the output waveform component associated with a waveform generator of a particular frequency. A phase shift calculator determines the phase shift between the input and output waveform components associated with the waveform generator. A phase shift to capacitance translator determines the instantaneous capacitance of the capacitive touch element as a function of the phase shift as described above. 
         [0016]    ADC and SB DFT-implemented embodiments may also include amplitude ratio logic to operate on the I and Q DFT outputs to determine the instantaneous amplitude of the output waveform component associated with a waveform generator of a particular frequency. An amplitude ratio calculator determines the ratio of the amplitude of the output waveform component associated with the waveform generator to the amplitude of the input waveform associated with that waveform generator. An amplitude ratio to capacitance translator determines the instantaneous capacitance of the capacitive touch element as a function of the amplitude ratio as described above. 
       DETAILED DESCRIPTION 
       [0017]      FIG. 1  is a schematic diagram of a capacitive touch detection apparatus  100  according to various example embodiments of the invention. The apparatus  100  includes a periodic waveform generator  105  coupled across a series-connected resistive-capacitive (“RC”) network  110 . The periodic waveform generator  105  generates a periodic voltage waveform of frequency F and imposes the periodic voltage waveform across the RC network  110 . The periodic waveform generator  105  may generate a sine waveform, a triangular waveform, a square waveform or a pulse waveform among other waveform shapes. 
         [0018]    The RC network  110  includes a capacitive touch element C  112  of variable capacitance C_TOUCH and a resistor R  115 . Values of the capacitance parameter C_TOUCH are a function of a proximity of a dielectric mass to C  112 , among other factors. 
         [0019]    The detection apparatus  100  also includes a phase shift detector  120 . A first input  122  of the phase shift detector  120  is coupled across the RC network  110  to receive the periodic voltage waveform. A second input  124  of the phase detector  120  is communicatively coupled across either C  112  or R  115  to receive a decreased magnitude, phase-shifted version of the periodic voltage waveform. The phase shift detector  120  measures a phase shift φ_DIFF between the periodic voltage waveform and the decreased magnitude, phase-shifted version of the periodic voltage waveform. The decreased magnitude, phase-shifted version of the periodic voltage waveform is also referred to herein as “the output waveform.” It is noted that the output waveform, although shown in the figures as being sensed across C  112 , may alternatively be sensed across R  115 . 
         [0020]    The detection apparatus  100  further includes a phase shift to capacitance translator  130  coupled to an output  132  of the phase shift detector  120 . The phase shift to capacitance translator  130  converts φ_DIFF to a first measure of C_TOUCH as C_TOUCH(φ_DIFF)=(tan φ_DIFF)/(2*π*f*R). 
         [0021]    Some embodiments of the detection apparatus  100  also include an amplitude ratio detector  145 . A first input  148  of the amplitude ratio detector  145  is coupled across the RC network  110  to receive the periodic voltage waveform. A second input  152  is communicatively coupled across either C  112  or R  115  to receive the decreased magnitude, phase-shifted version of the periodic voltage waveform. The amplitude ratio detector  145  measures an amplitude of the periodic voltage waveform and an amplitude of the decreased magnitude, phase-shifted version of the periodic voltage waveform. From these two amplitude values, the amplitude ratio detector  145  determines a ratio (“A_RATIO”) of the amplitude of the decreased magnitude, phase-shifted version of the periodic voltage waveform to the amplitude of the periodic voltage waveform. 
         [0022]    Embodiments of the capacitive touch detection apparatus  100  which include the amplitude ratio detector  145  also include an amplitude ratio to capacitance translator  160 . The amplitude ratio to capacitance translator  160  is coupled to the amplitude ratio detector  145  to convert A_RATIO to a second measure of C_TOUCH, “C_TOUCH(A_RATIO)”. The amplitude ratio to capacitance translator  160  generates the second measure of C_TOUCH as C_TOUCH(A_RATIO)=[(1−A_RATIO**2)]**1/2/(2*π*f*R*A_RATIO). 
         [0023]    In some embodiments, the capacitive touch detection apparatus  100  includes an amplifier  135  input-coupled across either C  112  or R  115 , whichever the output waveform measurements are taken from. An output  138  of the amplifier  135  is coupled to one or more of the second input  124  of the phase shift detector  120  and/or the second input  152  of the amplitude ratio detector. The amplifier  135  reduces impedance loading of C  112  or R  115  by the phase shift detector  120 , the amplitude ratio detector  145 , or both. 
         [0024]    Some embodiments of the capacitive touch detection apparatus  100  include C_TOUCH validation logic  165 . The C_TOUCH validation logic  165  is coupled to the phase shift to capacitance translator  130  and to the amplitude ratio to capacitance translator  160  via inputs  168  and  170 , respectively. The C_TOUCH validation logic  165  receives one or more values of C_TOUCH(φ_DIFF) and/or one or more values of C_TOUCH(A_RATIO). Some embodiments of the C_TOUCH validation logic  165  determine one or more differences between the value(s) of C_TOUCH(φ_DIFF) and the value(s) of C_TOUCH(A_RATIO). The latter embodiments may determine whether the difference(s) are greater than a maximum selected difference (“C_TOUCH_MAX_DIFF”). Such embodiments of the C_TOUCH validation logic  165  may output a value of C_TOUCH termed “C_TOUCH_OUT” as a function of at least one of C_TOUCH(φ_DIFF) or C_TOUCH(A_RATIO) if the difference is less than or equal to C_TOUCH_MAX_DIFF. The latter embodiments may output an invalidity indication if the difference is greater than C_TOUCH_MAX_DIFF. Some embodiments of the capacitive touch detection apparatus  100  may also include a register  175  coupled to the C_TOUCH validation logic  165  to store the value of C_TOUCH_MAX_DIFF. 
         [0025]      FIG. 2  is a waveform diagram of an input periodic waveform  210  and a corresponding output periodic waveform  215  as generated by the various example embodiments disclosed herein. The output waveform  215  is a decreased magnitude, phase-shifted version of the input periodic waveform  210 . The various embodiments herein measure the phase difference φ_DIFF  220  between the input and output waveforms  210  and  215 , respectively. These embodiments determine instantaneous values of the variable capacitance C_TOUCH of a capacitive sensor as a function of φ_DIFF  220 . 
         [0026]    Some embodiments also measure the magnitudes A_INPUT  230  and A_OUTPUT  235  of the input and output periodic waveforms  210  and  215 , respectively. These embodiments determine instantaneous values of the variable capacitance C_TOUCH of a capacitive sensor as a function of a ratio of A_OUTPUT  235  to A_INPUT  230 . Although the measures of A_INPUT  230  and A_OUTPUT  235  are shown as peak-to-peak values, some embodiments may measure the magnitudes of the input and output waveforms  210  and  215  as zero-to-peak values or use other magnitude measurement points. As previously mentioned, although the periodic waveforms are shown in the example of  FIG. 2  as sine waves, periodic waveforms of any shape may be implemented by the various embodiments, including triangular waveforms, square waveforms, etc. 
         [0027]      FIGS. 3 and 4  are schematic diagrams of capacitive touch detection apparatus  300  and  400 , respectively, according to various example embodiments. The apparatus  300  and  400  will be described together to avoid repetitiveness as they are similar. The apparatus  300  processes touch sensor signals initiated by a single periodic waveform generator  305 A. The apparatus  400  performs the same processing techniques as the apparatus  300  on signals initiated by any number N of waveform generators. It is noted, however, that for the sake of clarity  FIG. 4  shows circuitry associated with two waveform generators  305 A and  305 B as representative of circuitry associated with N waveform generators. Thus, the third through Nth copy of schematic diagram elements and/or signal names associated with a number of waveform generators greater than two referred to below are not shown on  FIG. 4 . The apparatus  300  and  400  will be referred to below either individually as “apparatus  300 ” and “apparatus  400 ” or collectively as “apparatus  300 / 400 .” 
         [0028]    The capacitive touch detection apparatus  300 / 400  includes one or more periodic waveform generators (e.g., the waveform generators  305 A and  305 B) coupled across a series-connected RC network  110 . The RC network  110  includes a capacitive touch element C  112  of variable capacitance C_TOUCH and a resistor R  115 . Values of C_TOUCH are a function of a proximity of a dielectric mass to C  112 . Each of the periodic waveform generator(s)  305 A,  305 B . . . periodic waveform generator N generates a periodic voltage waveform of frequency F1, F2 . . . or FN of a known amplitude (“A_INPUT_F1”, “A_INPUT_F2” . . . or “A_INPUT_FN”, respectively) and a known phase value (“φ_INPUT_F1”, “φ_INPUT_F2” . . . or “φ_INPUT_FN”, respectively). A sum (“F_SUM”) of the periodic voltage waveforms is imposed across the RC network  110 . In the case of the apparatus  300 , F_SUM is equivalent to the single input periodic waveform generated by the single waveform generator  305 A. 
         [0029]    The capacitive touch detection apparatus  300 / 400  also includes an ADC  308  communicatively coupled across either C  112  or R  115 . The ADC  308  receives and periodically samples a decreased magnitude, phase-shifted version of F_SUM appearing across C or R, whichever the output waveform measurements are taken from. The ADC  308  samples the decreased magnitude, phase-shifted version of F_SUM according to a first sampling periodicity associated with the periodic voltage waveform of frequency F1, a second sampling periodicity associated with the periodic voltage waveform of frequency F2 . . . and an Nth sampling periodicity associated with the periodic voltage waveform of frequency FN. The ADC  308  outputs a numerical magnitude value corresponding to each periodic sample. 
         [0030]    Some embodiments of the capacitive touch detection apparatus  300 / 400  also include one or more sample value table(s)  310 , each sample value table  310  coupled to the ADC  308 . It is noted that although sample value tables(s) are not shown in  FIG. 4 , some embodiments of the apparatus  400  may include sample value tables as described herein. Each sample value table stores a corresponding set of numerical magnitude values sampled according to the first sampling periodicity, the second sampling periodicity . . . or the Nth sampling periodicity. 
         [0031]    The capacitive touch detection apparatus  300 / 400  also includes DFT logic modules DFT F1  312 A, DFT F2  312 B . . . DFT FN communicatively coupled to the ADC. Each DFT logic module  312 A,  312 B . . . DFT FN corresponds to a periodic waveform generator  305 A,  305 B . . . or periodic waveform generator FN. In some embodiments, the DFT logic modules  312 A,  312 B . . . DFT FN may be implemented as single bin (“SB”) DFTs. Each DFT logic module  312 A,  312 B . . . DFT FN performs DFT operations on a set of numerical magnitude values sampled from one or more portions of a cycle of the decreased magnitude, phase-shifted version of the waveform F_SUM. Each set of sample values corresponds to the first sampling periodicity, the second sampling periodicity . . . or the Nth sampling periodicity. Each DFT logic module  312 A,  312 B . . . DFT FN generates a real numerical output value (“I_OUT_F1”, “I_OUT_F2” . . . or “I_OUT_FN”) and an imaginary numerical output value (“Q_OUT_F1”, “Q_OUT_F2” . . . or “Q_OUT_FN”) for each set of numerical magnitude sample values. If included in an embodiment of the capacitive touch detection apparatus  300 / 400 , each sample value table  310  is coupled to a corresponding DFT logic module to store a set of numerical magnitude values as input values to the corresponding DFT logic module. 
         [0032]    Some embodiments of the capacitive touch detection apparatus  300 / 400  also include a sampling clock (e.g., the sampling clock  315  of  FIG. 3 ) coupled to the ADC  308  and communicatively coupled to the periodic waveform generators  305 A,  305 B . . . periodic waveform generator FN. The sampling clock (not shown on  FIG. 4 ) clocks the periodic samples into the ADC  308 . The sampling clock also synchronizes the periodic waveform generators  305 A,  305 B . . . periodic waveform generator FN to the sampling periodicities in order to provide accurate phase measurements between F_SUM and the decreased magnitude, phase-shifted version of F_SUM. 
         [0033]    Some embodiments of the capacitive touch detection apparatus  300 / 400  also include a clock divider circuit “÷M” (e.g., the ÷M circuit  318  of  FIG. 3 ). The ÷M circuit  318  lengthens the period of the sampling clock by an integer multiple M. Doing so by certain values of M such as the integer value four results in integer values in the sets of sample values generated by the ADC  308 . Receiving integer values may simplify DFT operations by limiting such operations to addition and subtraction and avoiding more processor-intensive multiplication and division operations. 
         [0034]    The capacitive touch detection apparatus  300 / 400  further includes one or more phase shift logic modules (e.g., the phase shift logic module  322  of  FIG. 3 ). It is noted that components of each phase shift logic module associated with the apparatus  400  are shown in  FIG. 4  and described below with reference to both  FIGS. 3 and 4 . However, for purposes of clarity, outlines and labels of the phase shift logic module blocks of the apparatus  400  are not shown in  FIG. 4 . Each phase shift logic module is coupled to a DFT logic module  312 A,  312 B . . . or DFT FN of a corresponding frequency. Each phase shift logic module receives the I_OUT value and the Q_OUT value from the corresponding DFT logic module  312 A,  312 B . . . or DFT FN. Each phase shift logic module determines a phase shift (“φ_DIFF_F1”, “φ_DIFF_F2” . . . or “φ_DIFF_FN”) between the periodic voltage waveform of frequency F1, F2 . . . or FN and the decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN associated with the corresponding DFT logic module. 
         [0035]    Each phase shift logic module of the apparatus  300 / 400  includes an output phase calculator (e.g., the F1 output phase calculator  325 A and the F2 output phase calculator  325 B). Each output phase calculator is coupled to the DFT_F1  312 A, DFT_F2  312 B . . . or to the DFT_FN corresponding to the frequency of the output phase calculator. For example, the F1 output phase calculator  325 A is coupled to the DFT F1  312 A. Each output phase calculator  325 A,  325 B . . . or FN output phase calculator determines an output phase value (“φ_OUTPUT_F1”, “φ_OUTPUT_F2” . . . or “φ_OUTPUT_FN”) associated with a corresponding decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN. For example, the F1 output phase calculator  325 A determines φ_OUTPUT_F1 as equal to arctan 2(I_OUT_F1,Q_OUT_F1). The F2 output phase calculator  325 B determines φ_OUTPUT_F2 as equal to arctan 2(I_OUT_F2,Q_OUT_F2). The FN output phase calculator determines φ_OUTPUT_FN as equal to arctan 2(I_OUT_FN,Q_OUT _FN). 
         [0036]    Each phase shift logic module of the apparatus  300 / 400  also includes an F1 input phase register  328 A, an F2 input phase register  328 B . . . or FN input phase register. Each input phase register stores a known input phase value φ_INPUT_F1, φ_INPUT F2 . . . or φ_INPUT_FN associated with the corresponding periodic voltage waveform of frequency F1, F2 . . . or FN. 
         [0037]    Each phase shift logic module of the apparatus  300 / 400  also includes a phase shift calculator (e.g., the F1 phase shift calculator  332 A and the F2 phase shift calculator  332 B). Each phase shift calculator is coupled to an output phase calculator of a corresponding frequency and to an input phase register of the corresponding frequency. For example, the F1 phase shift calculator  332 A is coupled to the F1 output phase calculator  325 A and to the F1 input phase register  328 A. Each phase shift calculator determines the phase shift φ_DIFF_F1, φ_DIFF_F2 . . . or φ_DIFF_FN corresponding to the input and output waveforms of frequencies F1, F2 . . . and FN, respectively. For example, the F1 phase shift calculator  332 A determines φ_DIFF_F1 as equal to (φ_OUTPUT_F1)−(φ_INPUT_F1). The F2 phase shift calculator  332 B determines φ_DIFF_F2 as equal to (φ_OUTPUT_F2)−(φ_INPUT_F2). The FN phase shift calculator determines φ_DIFF_FN as equal to (φ_OUTPUT_FN)−(φ_INPUT_FN). 
         [0038]    The capacitive touch detection apparatus  300 / 400  includes one or more phase shift to capacitance translators (e.g., the F1 phase shift to capacitance translator  130 A and the F2 phase shift to capacitance translator  130 B). Each phase shift to capacitance translator is coupled to a corresponding phase shift logic module (e.g., the phase shift logic module  322 ) of a corresponding frequency. Each phase shift to capacitance translator  130 A,  130 B . . . FN phase shift to capacitance translator determines a potential value of C_TOUCH termed “C_TOUCH(φ_DIFF_F1)”, “C_TOUCH(φ_DIFF_F2)” . . . or “C_TOUCH(φ_DIFF_FN)”. Each C_TOUCH potential value C TOUCH(φ_DIFF_F1), C_TOUCH(φ_DIFF_F2) . . . or C_TOUCH(φ_DIFF_FN) corresponds to the phase shift between the periodic voltage waveform of frequency F1, F2 . . . or FN and the corresponding decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN. For example, the F1 phase shift to capacitance translator  130 A determines C_TOUCH(φ_DIFF_F1) as equal to (tan φ_DIFF_F1)/(2*π*F1*R). The F2 phase shift to capacitance translator  130 B determines C_TOUCH(φ_DIFF_F2) as equal to (tan φ_DIFF_F2)/(2*π*F2*R). The FN phase shift to capacitance translator determines C_TOUCH(φ_DIFF_FN) as equal to (tan φ_DIFF_FN)/(2*π*FN*R). 
         [0039]    Some embodiments of the capacitive touch detection apparatus  300 / 400  also include one or more amplitude ratio logic modules (e.g., the amplitude ratio logic module  340  of  FIG. 3 . It is noted that components of each amplitude ratio logic module associated with the apparatus  400  are shown in  FIG. 4  and described below with reference to both  FIGS. 3 and 4 . However, for purposes of clarity, outlines and labels of the amplitude ratio logic module blocks of the apparatus  400  are not shown in  FIG. 4 . Each amplitude ratio logic module is coupled to a DFT logic module of a corresponding frequency (e.g., the F1 amplitude ratio logic module is coupled to the DFT F1 logic module  312 A and the F2 amplitude ratio logic module is coupled to the DFT F2 logic module  312 B). Each amplitude ratio logic module receives the I_OUT value and the Q_OUT value from the corresponding DFT logic module and determines an amplitude ratio (“AR_F1”, AR_F2” . . . or “AR_FN”) of the decreased magnitude, phase-shifted version of the periodic voltage waveform associated with the corresponding DFT logic module to the periodic voltage waveform associated with the corresponding DFT logic module. 
         [0040]    Each amplitude ratio logic module includes an output amplitude calculator (e.g., the F1 output amplitude calculator  344 A and the F2 output amplitude calculator  344 B). Each output amplitude calculator is coupled to a DFT logic module of a corresponding frequency. For example, the F1 output amplitude calculator  344 A is coupled to the DFT F1 logic module  312 A. Each output amplitude calculator determines an output amplitude value (“A_OUTPUT_F1”, “A_OUTPUT_F2” . . . or “A_OUTPUT_FN”) associated with a corresponding decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN. For example, the F1 output amplitude calculator  344 A calculates A_OUTPUT_F1 as sqrt[(I_OUT_F1)**2+(Q_OUT_F1)**2]. The F2 output amplitude calculator  344 B calculates A_OUTPUT_F2 as sqrt[(I_OUT_F2)**2+(Q_OUT_F2)**2]. The FN output amplitude calculator calculates A_OUTPUT_FN as sqrt[(I_OUT_FN)**2+(Q_OUT_FN)**2]. 
         [0041]    In some embodiments of the capacitive touch detection apparatus  300 / 400  which include amplitude ratio logic module(s), each amplitude ratio logic module also includes an input amplitude register (e.g., the F1 input amplitude register  348 A and the F2 input amplitude register  348 B). Each input amplitude register stores the constant input amplitude value A_INPUT_F1, A_INPUT_F2 . . . or A_INPUT_FN associated with the corresponding periodic voltage waveform of frequency F1, F2 . . . or FN. 
         [0042]    Each amplitude ratio logic module also includes an amplitude ratio calculator (e.g., the F1 amplitude ratio calculator  352 A and the F2 amplitude ratio calculator  352 B). Each amplitude ratio calculator is coupled to the an output amplitude calculator and to an input amplitude register of the corresponding frequency. For example, the F1 amplitude ratio calculator  352 A is coupled to the F1 output amplitude calculator  344 A and to the F1 input amplitude register  348 A. Each amplitude ratio calculator determines an amplitude ratio AR_F1, AR_F2 . . . or AR_FN of the corresponding decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN to the periodic voltage waveform of frequency F1, F2 . . . or FN. For example, the F1 amplitude ratio calculator  352 A determines AR_F1 as equal to A_OUTPUT_F1/A_INPUT_F1. The F2 amplitude ratio calculator  352 B determines AR_F2 as equal to A_OUTPUT_F2/A_INPUT_F2. The FN amplitude ratio calculator determines AR_FN as equal to A_OUTPUT_FN/A_INPUT_FN. 
         [0043]    Embodiments of the capacitive touch detection apparatus  300 / 400  which include amplitude ratio logic module(s) also include one or more amplitude ratio to capacitance translators (e.g., the F1 amplitude ratio to capacitance translator  160 A and the F2 amplitude ratio to capacitance translator  160 B). Each amplitude ratio to capacitance translator is coupled to an amplitude ratio logic module (e.g., the amplitude ratio logic module  340 ) of a corresponding frequency. For example, the F1 amplitude ratio to capacitance translator  160 A is coupled to the F1 amplitude ratio logic module. Each amplitude ratio logic module determines a potential C_TOUCH capacitance value “C_TOUCH(AR_F1)”, “C_TOUCH(AR_F2)” . . . or “C_TOUCH(AR_FN)” corresponding to an amplitude ratio of the corresponding decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN to the periodic voltage waveform of frequency F1, F2 . . . or FN. For example, the F1 amplitude ratio to capacitance translator  160 A determines C_TOUCH(AR_F1) as equal to [(1−AR_F1**2)]**½/(2*π*F1*R*AR_F1). The F2 amplitude ratio to capacitance translator  160 B determines C_TOUCH(AR_F2) as equal to [(1−AR_F2**2)]**½/(2*π*F2*R*AR_F2). The FN amplitude ratio to capacitance translator determines C_TOUCH(AR_FN) as equal to [(1−AR_FN**2)]**½/(2*π*FN*R*AR_FN). 
         [0044]    Some embodiments of the capacitive touch detection apparatus  300 / 400  include C_TOUCH validation logic  165  as described above with respect to the apparatus  100 . The C_TOUCH validation logic  165  is coupled to the phase shift to capacitance translator(s)  130 A,  130 B . . . and to the Nth phase shift to capacitance translator. The C_TOUCH validation logic  165  is also coupled to the amplitude ratio to capacitance translator(s)  160 A,  160 B . . . and to the Nth amplitude ratio to capacitance translator. The C_TOUCH validation logic  165  receives one or more values of C_TOUCH(φ_DIFF) and/or one or more values of C_TOUCH(AR). 
         [0045]    Some embodiments of the C_TOUCH validation logic  165  compare two or more of the potential C_TOUCH values C_TOUCH(φ_DIFF_F1), C_TOUCH(φ_DIFF_F2) . . . C_TOUCH(φ_DIFF_FN), C_TOUCH(AR_F1), C_TOUCH(AR_F2) . . . and C_TOUCH(AR_FN). The comparison operations determine whether a difference between the potential C_TOUCH values is greater than a selected value C_TOUCH_MAX_DIFFERENCE. These embodiments of the C_TOUCH validation logic  165  output a numerical average of the potential C_TOUCH values as C_TOUCH_OUT if the difference between the potential C_TOUCH values is not greater than the selected value C_TOUCH_MAX_DIFFERENCE. An invalidity flag C_TOUCH_MEASUREMENT_INVALID is output from the C_TOUCH validation logic  165  if the difference between the potential C_TOUCH values is greater than the selected value C_TOUCH_MAX_DIFFERENCE. Some embodiments of the capacitive touch detection apparatus  300 / 400  also include a register  175  coupled to the C_TOUCH validation logic  165 . The register  175  contains the selected value C_TOUCH_MAX_DIFFERENCE. 
         [0046]    Some embodiment of the capacitive touch detection apparatus  300 / 400  perform weighted averaging operations on the potential C_TOUCH values C_TOUCH(φ_DIFF_F1), C_TOUCH(φ_DIFF_F2) . . . C_TOUCH(φ_DIFF_FN), C_TOUCH(AR_F1), C_TOUCH(AR_F2) . . . and C_TOUCH(AR_FN). Such embodiments output a weighted average of the potential C_TOUCH values as C_TOUCH_OUT. Some such embodiments may output averages of the potential C_TOUCH values as C_TOUCH_OUT without regard to differences between the potential C_TOUCH values. 
         [0047]      FIG. 5  is a frequency plot illustrating a filtering effect associated with the capacitive touch detection apparatus  300 / 400  according to various example embodiments. The DFT embodiments illustrated by the example apparatus  300 / 400  have the advantage of creating a narrow-band filter  510  around the waveform generator frequency  515 . The narrow-band filter  510  excludes previously-discussed EMI frequency components (e.g., the EMI components  520  and  525 ) occurring outside of the narrow-band filter  510  associated with the waveform generator frequency  515 . However, the frequencies of the EMI components may quickly change position over time. 
         [0048]    The apparatus  400  may overcome this problem by implementing two or more periodic waveform generators, each of a different frequency. The apparatus  400  may use capacitance values determined from measurements utilizing a waveform generator frequency not subject to unacceptable levels of EMI frequency components at the time of a particular instantaneous touch element capacitance measurement. Some embodiments of the capacitive touch detection apparatus  300  may perform a similar function by changing the frequency of the single periodic waveform generator when capacitance measurements are deemed unreliable by the C_TOUCH validation logic  165 . However, a subsequent C_TOUCH measurement at a different frequency may be subject to one or more EMI frequency components having moved to the new waveform generator frequency by the time the subsequent C_TOUCH measurement is made. 
         [0049]      FIGS. 6A-6C  are a flow diagram illustrating an example method  600  of capacitive touch detection according to various example activities. The method  600  commences at block  605  with generating one or more periodic waveforms. Each periodic waveform is of a frequency F1, F2 . . . or FN, of a known phase value φ_INPUT_F1, φ_INPUT_F2 . . . or φ_INPUT_FN and of a known amplitude A_INPUT_F1, A_INPUT_F2 . . . or A_INPUT_FN. 
         [0050]    The method  600  continues at block  610  with imposing a sum (“F_SUM”) of the periodic voltage waveforms across a series-connected RC network. The RC network includes a capacitive touch element C of variable capacitance C_TOUCH and a resistor R. The parameter C_TOUCH is a function of a proximity of a dielectric mass to C. 
         [0051]    The method  600  includes periodically sampling a decreased magnitude, phase-shifted version of F_SUM appearing across either C or R, at block  615 . The decreased magnitude, phase-shifted version of F_SUM is sampled according to a first sampling periodicity associated with the input periodic voltage waveform of frequency F1, a second sampling periodicity associated with the input periodic voltage waveform of frequency F2 . . . and an Nth sampling periodicity associated with the input periodic voltage waveform of frequency FN. 
         [0052]    The method  600  also includes performing DFT operations on a set of numerical magnitude values sampled according to the first sampling periodicity, the second sampling periodicity . . . or the Nth sampling periodicity from one or more portions of a cycle of the decreased magnitude, phase-shifted version of F_SUM, at block  620 . The method  600  further includes generating a real numerical output value I_OUT_F1, I_OUT_F2 . . . or I_OUT_FN and an imaginary numerical output value Q_OUT_F1, Q_OUT_F2 . . . or Q_OUT_FN from the DFT operations for each set of numerical magnitude sample values, at block  625 . 
         [0053]    The method  600  continues at block  628  with determining an output phase value φ_OUTPUT_F1, φ_OUTPUT_F2 . . . or φ_OUTPUT_FN. The output phase value is associated with a corresponding decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN. The output phase value is calculated as φ_OUTPUT_F1=arctan 2(I_OUT_F1,Q_OUT_F1), φ_OUTPUT_F2=arctan 2(I_OUT_F2,Q_OUT_F2) . . . and φ_OUTPUT_FN=arctan 2(I_OUT_FN,Q_OUT_FN1), respectively. 
         [0054]    The method  600  also includes determining a phase shift φ_DIFF_F1, φ_DIFF_F2 . . . or φ_DIFF_FN between the periodic voltage waveform of frequency F1, F2 . . . or FN and the decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN, at block  633 . The phase shift is calculated as φ_DIFF_F1=(φ_OUTPUT_F1)−(φ_INPUT_F1), φ_DIFF_F2=(φ_OUTPUT_F2)−(φ_INPUT_F2), and φ_DIFF_FN=(φ_OUTPUT_FN)−(φ_INPUT_FN), respectively. 
         [0055]    The method  600  further includes determining a potential value of C_TOUCH(φ_DIFF_F1), C_TOUCH(φ_DIFF_F2) . . . or C_TOUCH(φ_DIFF_FN), at block  638 . The potential value of C_TOUCH corresponds to the phase shift between the periodic voltage waveform of frequency F1, F2 . . . or FN and the corresponding decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN. The potential values of C_TOUCH are calculated as C_TOUCH(φ_DIFF_F1)=(tan φ_DIFF_F 1 )/(2*π*F1*R), C_TOUCH(φ_DIFF_F2)=(tan φ_DIFF_F2)/(2*π*F2*R) . . . and C_TOUCH(φ_DIFF_FN)=(tan φ_DIFF_FN)/(2*π*FN*R), respectively. 
         [0056]    The method  600  continues at block  640  with determining an output amplitude value A_OUTPUT_F1, A_OUTPUT_F2 . . . or A_OUTPUT_FN associated with a corresponding decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN. The output amplitude values are calculated as A_OUTPUT_F1=sqrt[(I_OUT_F1)**2+(Q_OUT_F1)**2], A_OUTPUT_F2=sqrt[(I_OUT_F2)**2+(Q_OUT_F2)**2] . . . and A_OUTPUT_FN=sqrt[(I_OUT_FN)**2+(Q_OUT_FN)**2], respectively; 
         [0057]    The method  600  includes determining an amplitude ratio AR_F1, AR_F2 . . . or AR_FN of the corresponding decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN to the periodic voltage waveform of frequency F1, F2 . . . or FN. The amplitude ratios are calculated as AR_F1=A_OUTPUT_F1/A_INPUT_F1, AR_F2=A_OUTPUT_F2/A_INPUT_F2 . . . and AR_FN=A_OUTPUT_FN/A_INPUT_FN, respectively. 
         [0058]    The method  600  also includes determining a potential value of C_TOUCH, C_TOUCH(AR_F1), C_TOUCH(AR_F2) . . . or C_TOUCH(AR_FN), at block  650 . The potential value of C_TOUCH corresponds to the amplitude ratio of the decreased magnitude, phase-shifted version of the periodic voltage waveform of frequency F1, F2 . . . or FN to the periodic voltage waveform of frequency F1, F2 . . . or FN. The potential values of C_TOUCH are calculated as C_TOUCH(AR_F1)=[(1−AR_F1**2)]**½/(2*π*F1*R*AR_F1), C_TOUCH(AR_F2)=[(1−AR_F2**2)]**½/(2*π*F2*R*AR_F2) . . . and C_TOUCH(AR_FN)=[(1−AR_FN**2)]**½/(2*π*FN*R*AR_FN), respectively. 
         [0059]    The method  600  further includes comparing two or more of the potential values of C_TOUCH, C_TOUCH(φ_DIFF_F1), C_TOUCH(φ_DIFF_F2) . . . C_TOUCH(φ_DIFF_FN), C_TOUCH(AR_F1), C_TOUCH(AR_F2) . . . and C_TOUCH(AR_FN), at block  655 . The method  600  continues at block  660  with determining whether a difference between the potential values of C_TOUCH is greater than a selected value C_TOUCH_MAX_DIFFERENCE. The method  600  includes generating a numerical average of the potential values of C_TOUCH as C_TOUCH_OUT if a difference between the potential values of C_TOUCH is not greater than the selected value C_TOUCH_MAX_DIFFERENCE, at block  665 . The method  600  also includes generating an invalidity flag C_TOUCH_MEASUREMENT_INVALID if a difference between the potential values of C_TOUCH is greater than the selected value C_TOUCH_MAX_DIFFERENCE, at block  670 . 
         [0060]    In some example sequences, the method  600  includes performing weighted averaging operations on the C_TOUCH potential values C_TOUCH(φ_DIFF_F1), C_TOUCH(φ_DIFF_F2) . . . C_TOUCH(φ_DIFF_FN), C_TOUCH(AR_F1), C_TOUCH(AR_F2) . . . and C_TOUCH(AR_FN), at block  675 . The weighted averaging operations may be performed without regard to differences between the potential values of C_TOUCH. The method  600  may, in such case, include generating a weighted average of the potential values of C_TOUCH as C_TOUCH_OUT, at block  680 . 
         [0061]    Methods and apparatus described herein may be useful in applications other than capacitive touch sensor detection. The apparatus  100 ,  300 , and  400  and the method  600  are intended to provide a general understanding of the sequences of various methods and the structures of various embodiments. They are not intended to serve as complete descriptions of all elements and features of methods, apparatus and systems that might make use of these example sequences and structures. 
         [0062]    The various embodiments may be incorporated into semiconductor analog and digital circuits for use in receptacle power converters, electronic circuitry used in computers, communication and signal processing circuitry, single-processor or multi-processor modules, single or multiple embedded processors, multi-core processors, data switches, and application-specific modules including multi-layer, multi-chip modules, among others. Such apparatus and systems may further be included as sub-components within a variety of electronic systems such as robotics, medical devices (e.g., heart monitor, blood pressure monitor, etc.), motor vehicles, televisions, cellular telephones, personal computers (e.g., laptop computers, desktop computers, handheld computers, tablet computers, etc.), workstations, radios, video players, audio players (e.g., MP3 (Motion Picture Experts Group, Audio Layer 3) players), set top boxes, household appliances and others. 
         [0063]    Methods and structures disclosed herein sense and measure capacitance variations from capacitive touch sensors by measuring phase shifts between sensor input and output signals. Phase shifts and amplitude ratio variations may be measured via apparatus and methods including DFT operations that filter EMI frequency components from the touch sensor signals. Sensor capacitance variations obtained from combinations of multi-frequency sensor signals, sensor input/output phase shifts and input/output signal magnitude ratio measurements may be compared to determine their validity. These structures and methods improve the sensitivity of capacitive sensor systems, including the unexpected result that capacitance variations down to single-digit femto-farads may be reliably sensed. 
         [0064]    By way of illustration and not of limitation, the accompanying figures show specific aspects in which the subject matter may be practiced. It is noted that arrows at one or both ends of connecting lines are intended to show the general direction of electrical current flow, data flow, logic flow, etc. Connector line arrows are not intended to limit such flows to a particular direction such as to preclude any flow in an opposite direction. The aspects illustrated are described in sufficient detail to enable those skilled in the art to practice the teachings disclosed herein. Other aspects may be used and derived therefrom, such that structural and logical substitutions and changes may be made without departing from the scope of this disclosure. This Detailed Description, therefore, is not to be taken in a limiting sense. The breadth of various aspects is defined by the appended claims and the full range of equivalents to which such claims are entitled. 
         [0065]    Such aspects of the inventive subject matter may be referred to herein individually or collectively by the term “invention” merely for convenience and without intending to voluntarily limit this application to any single invention or inventive concept, if more than one is in fact disclosed. Thus, although specific aspects have been illustrated and described herein, any arrangement calculated to achieve the same purpose may be substituted for the specific aspects shown. This disclosure is intended to cover any and all adaptations or variations of various aspects. 
         [0066]    The Abstract of the Disclosure is provided to comply with 37 C.F.R. §1.72(b) requiring an abstract that will allow the reader to quickly ascertain the nature of the technical disclosure. It is submitted with the understanding that it will not be used to interpret or limit the scope or meaning of the claims. In the preceding Detailed Description, various features are grouped together in a single embodiment for the purpose of streamlining the disclosure. This method of disclosure is not to be interpreted to require more features than are expressly recited in each claim. Rather, inventive subject matter may be found in less than all features of a single disclosed embodiment. The following claims are hereby incorporated into the Detailed Description, with each claim standing on its own as a separate embodiment.