Abstract:
An interface circuit from Common Mode Logic to a low voltage, fixed common mode output, with high current drive. The CML signal is received, and then re-referenced to a low-voltage band-gap supply. The circuit is arranged to provide an output data signal referenced to a second positive reference voltage supply responsive to receipt of a common mode input data signal referenced to a first positive reference voltage supply. The circuit avoids use of vertical PNP transistors in the signal path.

Description:
FIELD OF THE INVENTION 
     The present invention relates to a method and apparatus for high speed level shifting to provide a low voltage output, and a system incorporating the same. 
     BACKGROUND TO THE INVENTION 
     Logic signals at very high speed are often processed using bipolar Current Mode Logic (CML)—a signalling system involving small voltage swings (typically 250 mV) referenced to the positive supply of the integrated circuit (IC). Output interface levels from an IC using this logic are generally specified with a greater signal swing than 250 mV. 
     For a bipolar transistor to operate at speed it is necessary that its collector base junction does not enter forward conduction. Therefore, in order to allow increased signal swing on the output stage, the logic signals which drive it have to be level shifted from the positive rail reference down to a lower voltage. The more the signal can be shifted the greater the potential output swing. 
     Known attempts to level shift very high speed logic signals from a positive power rail reference down to a level as close as possible to a ground or negative rail, however, have encountered a number of problems. 
     Use of complementary vertical PNP transistors in the signal path slowed the signal since PNP&#39;s are inherently slower than NPN&#39;s. Their use also added considerably to the complexity of the semiconductor process since extra process steps were required in their fabrication. 
     Use of NPN diode level shifters was also considered. However this approach did not take account of power supply variation and required circuit complexity to allow for the temperature variation of the forward voltage drop of the diode junction. Tolerancing these circuits over temperature and power rail variation was therefore difficult since, in general, complexity in the signal path leads to slow circuits. 
     OBJECT OF THE INVENTION 
     The present invention seeks to provide an improved method and apparatus for high speed level shifting for low voltage output. 
     SUMMARY OF THE INVENTION 
     According to a first aspect of the present invention there is provided an electronic circuit arranged to receive a common mode input data signal referenced to a first positive reference voltage supply and to provide an intermediate data signal referenced to a second positive reference voltage supply, wherein said second reference voltage is lower than said first reference voltage. 
     Preferably, all transistors through which data signals pass between said input data signal and said intermediate data signal are NPN transistors. 
     Advantageously, level shifting of very high speed logic signals from the positive power rail reference down to a level as close as possible to the ground power rail without use of vertical PNP transistors in the signal path avoids slowing the signal since PNP&#39;s are inherently slower than NPN&#39;s, and avoids adding considerable complexity of the semiconductor fabrication process that would arise from additional steps required to form vertical PNP&#39;s. 
     Advantageously, signals referenced to the ground rail may be used to drive a very fast long tailed pair output stage with minimal loss of voltage headroom and hence best efficiency. 
     Advantageously, the present invention maximises the drive capability of such a high speed output signal without invoking any additional power rails or any special process steps in the fabrication of the IC. 
     Advantageously, the circuit may operate from a single external power supply. 
     Preferably said input data signal, and intermediate data signal are common mode logic signals. 
     Preferably said first and second voltages are both positive. 
     Advantageously this does not require provisioning of a negative power rail which would be expensive, require more power, and impose greater voltage stress on the integrated chip. 
     Preferably the maximum voltage of said intermediate signal does not exceed said second reference voltage. 
     Preferably said second reference voltage is maintained within lower tolerances than said first reference voltage. 
     Preferably said second reference voltage is provided from a band gap supply. 
     Advantageously, the circuit takes account of power supply variation and overcomes the problems associated with temperature variation of the forward voltage dcp associated with use of diode junction: tolerancing these circuits over temperature and power rail variation is more straightforward since the circuitry involved is simpler. 
     In a preferred embodiment, said first circuit comprises a first and second input circuits, a sensing element, and a current mirror circuit. 
     In a preferred embodiment the circuit additionally comprises a second circuit arranged to regenerate said intermediate data signal whereby to provide an output data signal. 
     Preferably, said second circuit comprises a long-tailed pair. 
     The arrangement may also comprise a third circuit arranged to provide a second output data signal referenced to said second reference voltage supply and having high current drive. 
     Preferably said high current drive is drawn from said first reference voltage supply. 
     Preferably said third circuit comprises a feedback circuit, whereby to maintain said second output data signal within tolerances. 
     The arrangement may also comprise a laser modulator circuit arranged to provide an optical output data signal responsive to receipt of said output data signal. 
     According to a second aspect of the present invention there is provided a laser modulator package assembly comprising an electronic circuit arrangement according to the first aspect of the present invention. 
     Preferably said electronic circuit arrangement comprises a laser modulator circuit arranged to provide an optical output data signal responsive to receipt of said output data signal. 
     According to a third aspect of the present invention there is provided an optical transmitter circuit board comprising a laser modulator package. 
     According to a fourth aspect of the present invention there is provided a telecommunications system comprising an electronic circuit arrangement. 
     The invention also provides for a system for the purposes of digital signal processing which comprises one or more instances of apparatus embodying the present invention, together with other additional apparatus. 
     The invention is also directed to a method by which the described apparatus operates and including method steps for carrying out every function of the apparatus. 
     In particular, according to a fifth aspect of the present invention there is provided a method of level shifting an input data signal comprising the stages of providing said input data signal referenced to a first positive reference voltage supply, generating an intermediate data signal referenced to a second positive reference supply responsive to receipt of said input data signal, wherein said second reference voltage is lower than said first reference voltage. 
     In a preferred embodiment, the method additionally comprises the steps of generating an output data signal with high current drive and referenced to said second positive reference voltage supply with high current drive responsive to receipt of said intermediate data signal, 
     In a further preferred embodiment, the method additionally comprises the steps of providing an optical output data signal responsive to receipt of said intermediate data signal. 
     The preferred features may be combined as appropriate, as would be apparent to a skilled person, and may be combined with any of the aspects of the invention. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In order to show how the invention may be carried into effect, embodiments of the invention are now described below by way of example only and with reference to the accompanying figures in which: 
     FIG. 1 shows a first circuit in accordance with the present invention; 
     FIG. 2 shows a second circuit in accordance with the present invention arranged to drive a Low Voltage Differential Signal (LVDS) buffer; 
     FIG. 3 shows a second circuit in accordance with the present invention arranged to drive a laser modulator. 
    
    
     DETAILED DESCRIPTION OF INVENTION 
     Referring first to FIG. 1, there is shown a first embodiment of a high-speed level shift circuit for low voltage output. 
     The high speed level shift circuit  100  comprises a re-referencing stage  101  and a regenerating buffer stage  150 . In the arrangement shown, the level shift circuit is coupled to an output stage  160 . 
     The re-referencing stage  101  comprises first and second input stages  110 ,  120 , sensing element  130 , and current mirror  140 . 
     Input stage  110  comprises a connection to the chip power rail  111  (typically 3.3V±0.3) and a data signal input INPUTA and a signal output  112 . It comprises a NPN transistor Q 1 , resistor R 1  capacitor C 1 , and current source I 1 . The base connection of Q 1  is connected to the data input INPUTA; the collector of Q 1  is connected to the chip power rail  111 . Resistor R 1  is connected in series between the emitter of Q 1  and the output port  112 . Capacitor C 1  is connected in parallel across resistor R 1 . Current source I 1  is connected to the emitter of Q 1 . 
     Input stage  120  is similarly configured with corresponding elements Q 2 , R 2 , C 2 , and I 2 . 
     Turning now to the sensing element  130 , this comprises a resistor R 3  and PNP transistor Q 4 , resistor R 3  being connected in series between the chip power rail  111  and the emitter of Q 4 . The base of Q 4  is connected to low voltage power rail  131  whilst the collector of Q 4  is connected to the current mirror  140  via connection  132 . Q 4  may be a lateral PNP transistor. 
     The current mirror  140  comprises three NPN transistors Q 5 , Q 6 , and Q 7 . The base of each of these transistors is connected to output  132  of the sensing circuit  130  as is the collector of transistor Q 5 . Collectors of Q 6 , Q 7  are connected to outputs  122 ,  112  of input stages  120 ,  110  respectively. Emitters of each of the three transistors are connected to ground. 
     The regenerating buffer stage  150  comprises a long-tailed pair comprising two resistors R 4 , R 5 , two NPN transistors Q 8 , Q 9 , and a current supply  13 . The base of Q 8  is connected to the output  122  of input stage  120  and correspondingly the base of Q 9  is connected to output  112  of input stage  110 . The collector of Q 8  is connected via resistor R 4  to the low voltage power rail  131 ; similarly the collector of Q 9  is connected via resistor R 5  to the low voltage power rail  131 . The emitters of both Q 8  and Q 9  are connected via current supply R 3  to ground. Two output ports are provided  151 ,  152  connected to the collector of Q 9  and Q 8  respectively. 
     The output stage  160  comprises a pair of NPN transistors Q 10  and Q 11  and resistor R 6 . The bases of Q 10 , Q 11  are connected to ports  152 ,  151  respectively whilst the emitters are connected via resistor R 6  to ground. The collectors of Q 11 , Q 10  are connected to output ports OUTPUTA, OUTPUTB respectively. 
     The regenerating buffer stage  150  is used to set the minimum voltage required for the output stage  160  to operate and also sets the voltage across resistor R 6 . 
     Output stage  160  is used to set the voltage and power levels at the two outputs OUTPUTA and OUTPUTB. 
     In operation, an input signal is provided on pins INPUTA and INPUTB. This signal is typically a differential Current Mode Logic (CML) signal with a swing of about 250 mV with respect to the positive supply rail. This drives emitter followers Q 1  and Q 2  which are biased via the two current sources  11  and  12 . The emitters drive two resistors, R 1  and R 2 , the DC voltage drops across which are set by the current flowing through the mirror transistors Q 5 , Q 6 . The purpose of this voltage drop is to match the signal from the emitter followers into the long tailed pair formed by Q 8  and Q 9 . Capacitors C 1  and C 2  give a high frequency path past R 1  and R 2  to overcome the capacitive loading imposed by Q 8  and Q 9 . The voltage swing on the collectors of Q 8  and Q 9  is restricted to about 250 mV and with the feed from Q 1  and Q 2  both are fully switched. In order to prevent the saturation of Q 8  or Q 9  the drive to the bases should not exceed the positive power rail for Q 8  and Q 9 . This will still reverse the collector base junctions of Q 8  or Q 9  by 250 mV but this does not substantially impinge on operating speed and is the normal operating condition for CML circuits anyway. Examining the circuit it is apparent that the required current, I dc  through R 1  and R 2  is given by: 
     
       
         
           
             
               
                 
                   
                     I 
                     dc 
                   
                   = 
                   
                     
                       
                         ( 
                         
                           
                             V 
                             pos1 
                           
                           - 
                           
                             V 
                             pos2 
                           
                         
                         ) 
                       
                       - 
                       
                         V 
                         be 
                       
                     
                     R1 
                   
                 
               
               
                 
                   ( 
                   1 
                   ) 
                 
               
             
           
         
                 
         
             
         
      
     
     where V be , is the forward emitter base voltage of Q 1  or Q 2  V pos1  is the power rail  111  supply voltage, and V pos2  is the low voltage power rail  131  supply voltage. 
     The requirements on the drive to Q 8  and Q 9  are not particularly stringent. A match to within 50 mV of the ideal is sufficient and to a first approximation a PNP forward voltage drop may be matched to a NPN drop (i.e. with appropriate geometries the magnitude of their V be &#39;s may be considered the same for a given operating current). A low speed circuit formed by the slow lateral PNP transistor, Q 4 , and the resistor R 3  senses the difference in the potential ((V pos1 −V pos2 )−V be ). In the case where R 1 =R 2 =R 3  the low speed circuit sets up a current in the collector of Q 4 , exactly matching the required current in R 1  and R 2 . It is then a simple matter of mirroring this current via Q 5 , Q 6 , and Q 7  in order to set up the correct conditions on the bases of Q 8  and Q 9 . 
     In some applications it may be advantageous to add further circuitry to allow for the finite gain of the transistors and the effects of base width modulation with collector voltage (Early effect). Such techniques are well known in the art. 
     The drive requirements to the bases of Q 10  and Q 11  dictate the-choice of V pos2 . It is apparent from FIG. 1 that V pos2  programs the current steered by Q 10  and Q 11 . In the case where the base current drawn by Q 10  and Q 11  makes minimal impact on the voltage on Q 8  and Q 9  collectors respectively, the voltage, V 6 , across R 6  is given by equation (2). 
     
       
           V 6= V   pos2 −V be   (2) 
       
     
     where V be  is the voltage between base and emitter of transistor Q 10  or Q 1 . 
     The current, I, out through Q 10  and Q 11  collectors is given by equation (3). 
     
       
           I= ( V   pos2   −V   be )/ R 6  (3) 
       
     
     It is clearly possible to vary the current output by changing V pos2 . By assigning an appropriate temperature dependence to V pos2 , the current, I, may be made to have any required temperature characteristic. 
     The output voltage of Q 10  and Q 11  may safely swing down to a minimum, V min , given in equation (4). 
     
       
           V   min =( V   pos2   −V   be )+ V   sat   (4) 
       
     
     where V sat  is the minimum collector emitter voltage of Q 10  and Q 11  below which the transistors saturate and performance is degraded. 
     In summary, the re-referencing circuit  101  is arranged to receive a pair of common mode logic signals on its data inputs INPUTA, INPUTB referenced to the positive power rail  111  and re-references these inputs to a second reference voltage V pos2 . In addition, by virtue of its connection via Q 4  to the low voltage bandgap supply, the circuit ensures that the voltage of the output signals is normally less than that of the low-voltage supply  131 . This ensures that the intermediate output signals at  112 ,  122  have a constant common mode referenced to the ground rail and lower than the bandgap supply voltage, V pos2  on rail  131 . 
     The regenerating buffer circuit  150  is arranged to regenerate the intermediate output signals provided by the initial re-referencing circuit  101 , re-referenced to the low voltage bandgap supply, V pos2 , and ensures that the maximum voltage of the output signals does not exceed that of the low-voltage supply. 
     Referring now to FIG. 2, there is shown the design of an LVDS output driver, using the level shift circuit  100 . 
     This circuit comprises an additional control loop  210 , comprising four resistors R 7 -R 10 , amplifier AMP, and NPN transistor Q 12 . Resistors R 9 , R 10  are connected in parallel from the inverting input of the amplifier AMP to data outputs OUTPUTA and OUTPUTB respectively of output stage  160 . The non-inverting input of the amplifier is connected to a +1.2 V supply whilst its output is connected to the base of transistor Q 12 . The collector of transistor Q 12  is connected to chip power rail V pos1 . Resistors R 7 , R 8  are arranged in parallel to provide paths from the emitter of transistor Q 12  to OUTPUTB and OUTPUTA respectively. 
     Control loop  210  stabilises the mean output voltage to  1 . 2  v as required by the LVDS specification. This is provided by sensing the mean voltage of the outputs with two high value resistors R 9  and R 10  (where R 9 , R 10 &gt;&gt;R 7 , R 8 ) and comparing with a reference 1.2 V source in the amplifier, AMP which drives the emitter follower of Q 12 . V pos2  is set such that the output swing on nodes OUTPUTA and OUTPUTB is 300 mV (in the middle of the specification for output swing) when loaded with an external 100 ohm load. 
     If the current drawn by the bases of Q 10 , Q 11  should increase such that appreciable drops across R 4 , R 5  would result then it is possible to increase V pos2  by one V be  and use NPN emitter followers on the output of Q 8 /Q 9  collectors to overcome this problem. 
     When driving a laser diode it is normal to have standing DC current which sets the diode at its threshold, and an additional superposed pulse modulation. The amplitude of both these components varies widely from diode to diode—even of the same type and so these have to be programmed for each diode. 
     FIG. 3 shows the design of a pulse modulator for a laser diode D 1  driven from a 3.3V supply (V pos1 ) with variable modulation current. The arrangement comprises a high speed level shift circuit  100  and a laser diode output stage  300 . 
     The laser diode output stage comprises a laser diode D 1 , five NPN transistors Q 10 -Q 11 , Q 13 - 15  resistor R 11 , and two current sources  14 ,  15 . 
     The bases of Q 13 , Q 14  are coupled to data outputs  151 ,  152  respectively; collectors of Q 13 , Q 14  are coupled to the chip power rail  111 ; and emitters coupled to ground via power supplies  14 , 15  respectively. 
     The bases of Q 10 , Q 11  are coupled to the emitters of Q 13 , Q 14  respectively; emitters of both Q 10  and Q 11  are coupled to the collector of Q 12 , the emitter of which is coupled via resistor R 11  to ground. 
     The collector of Q 10  is coupled to the chip power rail  111 ; the collector of Q 11  which carries the regenerated input signal from INPUTA, is coupled to ground via laser diode D 1 . 
     In this arrangement emitter followers Q 13 , Q 14  are placed between R 4 , R 5  and Q 10 , Q 11  respectively, as described above. This allows the driving of higher current laser diodes, but is not mandatory for more sensitive laser diodes. The current through the output stage in this arrangement is set by a programmable current source, comprising NPN transistor Q 15  and resistor R 11 . The voltage V BIAS  on Q 15  base sets the output current and may be adjusted for each laser diode. The only performance penalty of this arrangement over that in FIG. 1 is the additional headroom needed to accommodate the current source. Effectively this is the saturation voltage, V sat , of Q 15 . 
     This circuit arrangement facilitates collocation of the driver circuit with the laser diode itself in the same package. If the driver is mounted exceptionally close to the laser the electrical interface to the laser can be kept simple thereby saving power in the laser driver. This power saving is certainly necessary if they are to be collocated: lasers work best when cool. By saving power in the driver it is possible to mount the driver close to the laser without needing to use, for example, Peltier cooling devices to refrigerate the whole assembly. 
     The power saving comes about because of the avoidance of the difficulty of driving fast signals to a laser over any distance. The reason is the low impedance presented by the laser diode: typically a few ohms. To drive fast signals over any distance it is necessary to use a transmission line terminated in the natural impedance of the transmission line. Typically, the impedance of a transmission line is 50 ohms although lower impedances may be made (though not down to the impedance of the diode). The diode end of the transmission line has to match the line impedance. This is done by adding resistance in series with the diode to raise the resultant impedance to that of the line. This series resistor is where the power is wasted and the additional power has to come from the laser driver. Typically, commercial high speed test gear and cabling is built for 50 ohms, thus if laser and driver are to be supplied as separate modules this is very much the preferred impedance since users can hook the two together and/or evaluate them individually using standard equipment. The cabling, and special high frequency connectors would all add cost to the arrangement. The increased power required from the laser driver needs a separate integrated circuit, usually in the very expensive Gallium Arsenide process, since that is better suited than silicon to such an increased level of output. The additional electrical power has to be supplied to the laser system and the resultant increase in heat dissipated. And, of course, one module takes less space than two linked by cable with connectors on them. 
     Furthermore, purchasers of such apparatus would prefer to purchase off-the-shelf circuits arranged to accept standard logic signal in, and give optical signals out, rather that be left the task of coupling lasers to their drivers themselves. 
     Any range or device value given herein may be extended or altered without losing the effect sought, as will be apparent to the skilled person for an understanding of the teachings herein.