Abstract:
An Electrically Erasable Programmable Read Only Memory (EEPROM) memory array (FIGS.  7  and  8 ) is disclosed. The memory array includes a plurality of memory cells arranged in rows and columns. Each memory cell has a switch ( 806 ) coupled to receive a first program voltage (PGMDATA) and a first select signal (ROWSEL). A voltage divider ( 804 ) is coupled in series with the switch. A sense transistor ( 152 ) has a sense control terminal ( 156 ) and a current path coupled between an output terminal ( 108 ) and a reference terminal ( 110 ). A first capacitor ( 154 ) has a first terminal coupled to the switch and a second terminal coupled to the sense control terminal. An access transistor ( 716 ) has a control terminal coupled to receive a read signal ( 721 ), and a current path coupled between the output terminal and a bit line ( 718 ).

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     Cross-reference is made to commonly assigned patent application Ser. No. 12/804,439, (TI-67911) entitled “ARRAY ARCHITECTURE FOR REDUCED VOLTAGE, LOW POWER, SINGLE POLY EEPROM” filed Jul. 20, 2010, the teachings of which are incorporated herein by reference in their entirety. 
     BACKGROUND OF THE INVENTION 
     Embodiments of the present invention relate to an Electrically Erasable Programmable Read Only Memory (EEPROM) array architecture for a cell having a single polycrystalline silicon gate. 
     Contemporary semiconductor integrated circuits typically perform much more complex functions than previous designs. Mixed mode circuits performing combined analog, digital, and memory functions are common for many applications. At the same time these mixed mode circuits must keep the manufacturing process as simple as possible to reduce cost and improve the process yield. A single polycrystalline silicon EEPROM cell of the prior art that may be compatible with existing complementary metal oxide silicon (CMOS) processes is illustrated at  FIG. 10 . The cell includes complementary floating gates  1012  and  1014  which serve as control gates for respective sense transistors. During a read operation, these sense transistors are accessed by read select transistors which connect the sense transistors to bit line (BL) and complementary bit line (BL_) terminals. Each cell includes a control circuit  1020  which receives global address and control signals and produces local control signals for the respective cell. Programming is accomplished, for example, by driving WR 1 _low, WR_EN 1  high, and WR_EN 2  low. In this state, N-channel transistor  1008  is on and N-channel transistor  1010  is off. Reference transistors  1004  and  1006  couple low and high signals between respective P-channel and N-channel transistors. Responsively, P-channel transistor  1002  is on and P-channel transistor  1000  is off. This programs positive charge on floating gate  1012  and negative charge on floating gate  1014 . One disadvantage of this cell is that it requires a separate control circuit  1020  for each cell. Another disadvantage is that it requires substantial layout area for the complementary floating gates  1012  and  1014 . Yet another disadvantage of this cell is that transistors  1004  through  1010  are constructed as large drain-extended transistors indicated by asterisks to preclude punch through at relatively high drain-to-source voltages. 
     Other single polycrystalline silicon EEPROM cells of the prior art may be manufactured together with analog and digital circuits on a single integrated circuit. Such EEPROM cells permit nonvolatile memory to be formed in mixed mode circuits for many applications. Chi et al. (U.S. Pat. No. 5,940,324) and Chen et al. (U.S. Pat. No. 6,930,002) both developed single polycrystalline silicon EEPROM cells that are programmed by band-to-band tunneling. The present inventors have developed an improved array architecture for a single polycrystalline silicon EEPROM cell that offers several advantages over single polycrystalline silicon memory cells of the prior art as will become apparent in the following discussion. 
     BRIEF SUMMARY OF THE INVENTION 
     In a preferred embodiment of the present invention, an Electrically Erasable Programmable Read Only Memory (EEPROM) array is disclosed. The memory array includes a plurality of memory cells arranged in rows and columns. Each memory cell has a switch coupled to receive a first program voltage and a control terminal coupled to receive a first select signal. A voltage divider is coupled in series with the switch. A sense transistor having a sense control terminal has a current path coupled between an output terminal and a reference terminal. A first capacitor has a first terminal coupled to the first switch and a second terminal coupled to the sense control terminal. An access transistor having a control terminal coupled to receive a read signal has a current path coupled between the output terminal and a bit line. 
    
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING 
         FIG. 1  is a top view of a single polycrystalline silicon gate EEPROM cell that may be used with the present invention; 
         FIG. 2  is a cross sectional view of the EEPROM cell of  FIG. 1  at the plane A-A′; 
         FIG. 3  is a schematic diagram showing programming a logical zero (program) in the EEPROM cell; 
         FIG. 4  is a schematic diagram showing programming of a logical one (erase) in the EEPROM cell; 
         FIG. 5A  is a schematic diagram showing stress on an unselected EEPROM cell storing a logical one for V CG =0 V and V TG =−5 V; 
         FIG. 5B  is a schematic diagram showing stress on an unselected EEPROM cell storing a logical one for V CG =+5 V V TG =0 V; 
         FIG. 5C  is a schematic diagram showing stress on an unselected EEPROM cell storing a logical zero for V CG =0 V V TG =+5 V; 
         FIG. 5D  is a schematic diagram showing stress on an unselected EEPROM cell storing a logical zero for V CG =−5 V V TG =0 V; 
         FIG. 6  is a schematic diagram of an EEPROM cell that may be used with an embodiment of the array architecture of the present invention; 
         FIG. 7  is a schematic diagram of an embodiment of the array architecture of the present invention; 
         FIG. 8A  is a schematic diagram of another EEPROM cell that may be used with another embodiment of the array architecture of the present invention; 
         FIG. 8B  is a program/erase timing diagram illustrating operation of the cell of  FIG. 8A ; 
         FIG. 9A  is a drive circuit that may be used with the memory cell of  FIG. 8A  to produce the equalization signal (EQ); 
         FIG. 9B  is a drive circuit that may be used with the memory cell of  FIG. 8A  to produce the program data (PGMDATA) and complementary program data (PGMDATA_) signals; 
         FIG. 9C  is another drive circuit that may be used with the memory cell of  FIG. 8A  to produce the program data (PGMDATA) and complementary program data (PGMDATA_) signals; and 
         FIG. 10  is a single polycrystalline EEPROM memory cell of the prior art. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Preferred embodiments of the present invention provide significant advantages over previous memory array architectures using single polycrystalline silicon EEPROM memory cells as will become evident from the following detailed description. The present inventors have disclosed a single polycrystalline silicon EEPROM cell in U.S. patent application Ser. No. 12/462,076, (TI-66531), filed Jul. 28, 2009, and incorporated herein by reference in its entirety. The following discussion briefly describes that EEPROM memory cell to provide a more complete understanding of the present invention. In the following discussion, P and N are used to indicate semiconductor conductivity type. A “+” or “−” sign after the to conductivity type indicates a relatively high or low doping concentration, respectively, of the semiconductor region. Furthermore, the same reference numerals are used in the drawing figures to indicate common circuit elements. 
     Referring to  FIG. 1 , there is a top view of a single polycrystalline silicon gate (Poly) EEPROM memory cell that may be used with the present invention. The cell includes N− isolation regions  120  and  126 . These N− isolation regions serve to electrically isolate P− well regions  160  and  162 , respectively, from a P type substrate. In operation, they are preferably biased to a positive supply voltage at terminals  100  and  102 . A control gate terminal  104  contacts P+ region  140  as well as N+ region  122 , both of which are formed within P− well region  160 . A tunnel gate terminal  106  contacts P+ region  142  as well as N+ region  130 , both of which are formed within P− well region  162 . A single polycrystalline silicon gate layer  156  overlies a part of both P− well regions and is self aligned with N+ regions  122  and  130 . An N-channel sense transistor is formed between the P− well regions  160  and  162 . The sense transistor includes drain terminal  108 , source terminal  110 , and control gate  152 . The sense transistor operates to indicate the data state of the polycrystalline silicon gate layer  156  as will be explained in detail. 
     The polycrystalline silicon gate layer  156  is often referred to as a floating gate, since it is only capacitively coupled and not directly connected to other elements of the memory cell. The polycrystalline silicon gate forms one terminal of a control gate capacitor  150  as well as one terminal of a tunnel gate capacitor  154 . Referring now to  FIG. 2 , there is a cross sectional view of the EEPROM cell of  FIG. 1  at the plane A-A′. An N+ buried layer  202  together with N− isolation region  120  electrically isolates P− well region  160  from P substrate  210 . Likewise, another N+ buried layer  204  together with N− isolation region  126  electrically isolates P− well region  162  from P substrate  210 . Shallow trench isolation regions  200  isolate active regions such as control gate capacitor  150 , sense transistor  152 , and tunnel gate capacitor  154 . An upper plate of the control gate capacitor is formed by a first part of polycrystalline silicon gate layer  156 . A lower plate of the control gate capacitor is formed adjacent the upper plate by P− well region  160 . The upper and lower plates are separated by a dielectric region to form the control gate capacitor  150 . In a similar manner, an upper plate of the tunnel gate capacitor  154  is formed by a second part of polycrystalline silicon gate layer  156 . A lower plate of the tunnel gate capacitor  154  is formed adjacent the upper plate by P− well region  162 . The upper and lower plates are separated by a dielectric region to form the tunnel gate capacitor  154 . 
     Referring now to  FIGS. 3 and 4 , a programming operation of the control gate layer of the memory cell will be explained in detail. Numeric voltage values in the following discussion and throughout the instant specification are given by way of example for the purpose of illustration and may vary with different manufacturing processes.  FIG. 3  is a schematic diagram of the memory cell of  FIGS. 1-2 . N− isolation regions  120  and  126  as well as N+ buried layers  202  and  204  are biased at 5 V throughout the operation. A 5 V signal is applied to control gate terminal  104 . P+ region  140  is electrically connected to P− well region  160 . Thus, P− well region  160  is also at 5 V. The capacitance of control gate capacitor  150  (C CG ) is much larger than the total capacitance (C T ) of tunnel gate capacitor  154 , sense transistor gate  152  (C XTR ), and associated parasitic capacitance. The coupling ratio C CG /(C CG +C T +C XTR ) is at least 0.8 and preferably 0.9 or greater. The polycrystalline silicon gate layer voltage, therefore, is approximately 4 V to 4.5 V. 
     A −5 V signal is also applied to the tunnel gate terminal  106 . P+ region  142  is electrically connected to P− well region  162  which is, therefore, also at −5 V. An inversion layer is formed adjacent a second part of polycrystalline silicon gate layer  156  at the tunnel gate capacitor  154  below the intervening dielectric region. This dielectric region is preferably silicon dioxide or other suitable dielectric material as is known in the art. N+ region  130  provides a source of electrons for the inversion layer and remains in conductive contact with the inversion layer. Thus, a high electric field is generated across the relatively thin dielectric region sufficient to induce Fowler-Nordheim tunneling of electrons from the inversion layer to the polycrystalline silicon gate layer  156 . This relatively higher concentration of electrons significantly increases the threshold voltage of sense transistor  152  and renders it nonconductive in a subsequent read operation. 
     This EEPROM memory cell offers several advantages over memory cells of the prior art. First, the critical electric field necessary for Fowler-Nordheim tunneling is developed by positive and negative voltages of comparable magnitudes. This avoids the need to generate a high voltage power supply or to incorporate special high voltage transistors in the manufacturing process. Second, programming by Fowler-Nordheim tunneling greatly reduces the power requirement compared to prior art hot carrier generation methods such as avalanche multiplication and band-to-band tunneling. Third, Fowler-Nordheim tunneling from the inversion layer to the polycrystalline silicon gate layer  156  provides uniform current density over the entire area of the tunnel gate capacitor  154 . Thus, current density is much less than with methods of the prior art where current flow was through a much smaller area. Such areas were edge-dependent and determined by overlapping gate and underlying implant regions. The reduced programming current density of the present invention greatly increases program/erase cycles and corresponding reliability of the memory cell. 
     Referring now to  FIG. 4 , an erase operation of the control gate layer of the memory cell will be explained in detail.  FIG. 4  is a schematic diagram of the memory cell of  FIGS. 1-2 . As previously discussed, N− isolation regions  120  and  126  as well as N+buried layers  202  and  204  are biased at 5 V throughout the operation. A −5 V signal is applied to control gate terminal  104 . P+ region  140  is electrically connected to P− well region  160 . Thus, P− well region  160  is also at −5 V. Due to the coupling ratio of control gate capacitor  150  (C CG ) and the total capacitance (C T ) of tunnel gate capacitor  154 , sense transistor gate  152 , and associated parasitic capacitance the polycrystalline silicon gate layer voltage is approximately −4 V to −4.5 V. The voltage difference across control gate capacitor  150  forms an inversion layer adjacent a first part of polycrystalline silicon gate layer  156  below the intervening dielectric region. The inversion layer is electrically connected to N+ region  122  and, therefore, maintains the high coupling ratio between C CG  and C T . 
     A 5 V signal is also applied to the tunnel gate terminal  106 . P+ region  142  is electrically connected to P− well region  162  which is, therefore, also at 5 V. The voltage difference between the polycrystalline silicon gate  156  and the P− well region  162  forms an accumulation region at the lower plate (P− well region  162 ) of tunnel gate capacitor  154 . The resulting high electric field generated across the relatively thin dielectric region is sufficient to induce Fowler-Nordheim tunneling of electrons from polycrystalline silicon gate layer  156  to the accumulation region. Thus, a relatively lower concentration of electrons significantly decreases the threshold voltage of sense transistor  152  and renders it conductive in a subsequent read operation. 
     The previously discussed advantages of the EEPROM memory cell are also present during an erase operation. The critical electric field necessary for Fowler-Nordheim tunneling is developed by positive and negative voltages of comparable magnitudes. This avoids the need to generate a high voltage power supply or to incorporate special high voltage transistors in the manufacturing process. Programming by Fowler-Nordheim tunneling greatly reduces the power requirement compared to prior art hot carrier generation methods such as avalanche multiplication and band-to-band tunneling. Finally, Fowler-Nordheim tunneling from the polycrystalline silicon gate layer  156  to the accumulation region provides uniform current density over the entire area of the tunnel gate capacitor  154 . Thus, current density is much less than with methods of the prior art where current flow was through a much smaller area. Such areas were edge-dependent and determined by overlapping gate and underlying implant regions. The reduced programming current density of the present invention greatly increases program/erase cycles and corresponding reliability of the memory cell. 
     Turning now to  FIGS. 5A-5D , stress on unselected memory cells as in  FIGS. 1-2  of a memory array during programming of selected memory cells will be discussed in detail. Voltage stress on these unselected memory cells is due to the coupling ratio as previously discussed with regard to  FIGS. 3 and 4 . In the following discussion it should be understood that this stress may degrade data stored on the unselected memory cells after many programming (or erase) operations are performed on nearby selected memory cells. In particular,  FIG. 5A  is a schematic diagram showing stress on an unselected EEPROM cell storing a logical one for V CG =0 V and V TG =−5 V. By way of example, the floating gate voltage (V FG ) for a logical one is 4 V. When V TG =−5 V for programming a selected memory cell, the unselected memory cell of  FIG. 5A  has approximately −8.5 V across tunnel gate capacitor  154 . This stress causes positive charge loss  500  over many programming or erase operations, which greatly reduces the number of memory program/erase cycles and corresponding reliability of the memory cell. 
     Referring to  FIG. 5B , there is a schematic diagram showing stress on an unselected EEPROM cell storing a logical one for V CG =+5 V and V TG =0 V. As previously discussed, the floating gate voltage (V FG ) for a logical one is 4 V. When V CG =+5 V for programming a selected memory cell, the unselected memory cell of  FIG. 5B  has approximately −8.0 V across tunnel gate capacitor  154 . This stress will also cause positive charge loss  502  over many programming or erase operations. 
     Referring next to  FIG. 5C , there is a schematic diagram showing stress on an unselected EEPROM cell storing a logical zero for V CG =0 V and V TG =+5 V. Here, however, the floating gate voltage (V FG ) for a logical zero is −4 V. When V TG =+5 V for programming a selected memory cell, the unselected memory cell of  FIG. 5C  again has approximately +8.5 V across tunnel gate capacitor  154 . This stress causes negative charge loss  504  over many programming or erase operations, which greatly reduces the number of memory program/erase cycles and corresponding reliability of the memory cell. 
     Finally, referring to  FIG. 5D , there is a schematic diagram showing stress on an unselected EEPROM cell storing a logical zero for V CG =−5 V and V TG =0 V. As previously discussed, the floating gate voltage (V FG ) for a logical zero is −4 V. When V CG =−5 V for programming a selected memory cell, the unselected memory cell of  FIG. 5D  has approximately 9 V across tunnel gate capacitor  154 . This stress will also cause negative charge loss  506  over many programming or erase operations. 
     Turning now to  FIG. 6 , there is a schematic diagram of an EEPROM memory cell with surrounding circuitry that forms an element of the array architecture of the present invention. Recall from the previous discussion regarding  FIGS. 5A-5D  that stress on unselected memory cells occurs when a selected memory cell on the same tunnel gate lead or the same control gate lead is programmed. This stress depends on the voltage applied to the tunnel gate lead or control gate lead as well as the data state of the unselected memory cell. According to the present invention, program data lead  606  is selectively connected to tunnel gate lead  106  by switch  602 . Likewise, complementary program data lead  608  is selectively connected to control gate lead  104  by switch  604 . Both switches  602  and  604  are controlled by row select signal (ROWSEL) applied to lead  600 . Both program data leads  606  and  608  are generally perpendicular to the row select signal in the memory array. Only a selected cell, therefore, will have programming voltages applied to leads  606  and  608  when switches  602  and  604  are turned on by an active row select signal on lead  600 . This advantageously eliminates any stress to unselected memory cells that might degrade stored data states. 
     Referring now to  FIG. 7 , there is a schematic diagram of an embodiment of the array architecture of the present invention. For the purpose of illustration, the memory array includes selected memory cells  730  and  740 , which are already programmed to logical zero and one, respectively. The memory array also includes unselected memory cells  750 ,  760 , and  770 . Memory cells  730  and  740  are connected to row select leads  700  and  702 , which are oriented horizontally through the memory array. Memory cell  730  is connected to program data lines  704  and  706  via switches  712  and  714 , respectively. Program data lines  704  and  706  are oriented vertically through the memory array and generally perpendicular to row select leads  700  and  702 . Memory cell  740  is connected to program data lines  708  and  710 , via switches  722  and  724 , respectively. Program data lines  708  and  710  are also oriented vertically through the memory array and generally perpendicular to row select leads  700  and  702 . Finally, memory cells  730  and  740  include respective access transistors  716  and  726  to couple their stored data states to respective read bit leads  718  and  728 . 
     Unselected memory cells  750  and  760  share the same program data leads as selected memory cells  730  and  740 , respectively. The switches of unselected memory cells  750  and  760 , however, share different row select leads from selected memory cells  730  and  740 . Thus, the switches of unselected memory cells remain off when selected memory cells  730  and  740  are programmed and are not stressed as previously described with regard to  FIGS. 5A-5D . Moreover, the control gate and tunnel gate leads of memory cell  750  are connected to ground or a suitable reference voltage by equalization transistors  752  and  754 . Likewise, the control gate and tunnel gate leads of memory cell  760  are connected to ground or the suitable reference voltage by equalization transistors  762  and  764 . Unselected memory cells  750  and  760 , therefore, are not stressed and their respective data states remain intact when memory cells  730  and  740  are programmed. 
     Unselected memory cell  770  shares the same row select leads as selected memory cells  730  and  740 . The switches of unselected memory cell  770 , therefore, are on when the switches of selected memory cells  730  and  740  are on. The program data leads of unselected memory cell  770 , however, remain at zero volts or a suitable reference voltage. The control gate and tunnel gate leads of memory cell  770 , therefore, are not stressed as previously described with regard to  FIGS. 5A-5D . 
     In a first embodiment of  FIG. 7 , the switches of each memory cell are formed from complementary metal oxide semiconductor (CMOS) pass gates. Each CMOS pass gate is formed from an N-channel transistor in parallel with a P-channel transistor. Furthermore, in this first embodiment of the present invention, the voltage swing of the control gates of the switches is the same as the voltage swing on the program data leads (+V P  to −V P ), so that the switches of unselected cells are completely off when selected memory cells in the same column are programmed. 
     The maximum voltage across the control gate dielectric of the N-channel and P-channel transistors is generally the same as the programming voltage across the tunnel gate dielectric. This may be acceptable in some applications where programming time of the memory cells is not critical and some Fowler-Nordheim tunneling through the switch transistors is acceptable. In a second embodiment of the present invention, the switch transistors are separately ion implanted to preferentially grow a slightly thicker gate dielectric than that of the tunnel gate capacitors. In this second embodiment, programming voltage across tunnel gate capacitors may be safely increased and programming time decreased without damage to the switch transistors. 
     Turning now to  FIG. 8A , there is a modified memory cell that may be used in a third embodiment of the memory array of  FIG. 7 . The modified memory cell of  FIG. 8A  differs from the previously described memory cells of  FIG. 7  in three respects. First; each CMOS pass gate or switch now includes series-connected voltage divider transistors such as transistors  800  and  804  as well as switching transistors  802  and  806 . Second, row select signal ROWSEL operates at a reduced voltage swing of 0V to 5V (+V p ). Complementary row select signal ROWSEL_ operates at a reduced voltage swing of 0V to −5V (−V P ). Third, N-channel transistors  811  and  813  are added to the equalization circuit to hold control gate lead  104  and tunnel gate lead  106  to ground (0 V) when the memory cell is unselected. 
     Operation of the modified memory cell of  FIG. 8A  will now be explained in detail with reference to the program/erase timing diagram of  FIG. 8B . The left half of the timing diagram ( FIG. 8B ) illustrates operation when the memory cell is on a selected row. The memory cell row is selected at time t 0  when ROWSEL is high (0 V), ROWSEL_ is low (0 V), and EQ is low (−5 V). In this case, leads TG  106  and CG  104  are driven to −V TN  as illustrated by voltage levels  830  and  840 , respectively, by N-channel transistors of the CMOS switches. Here and in the following discussion, V TN  refers to a threshold voltage of an N-channel transistor and V TP  refers to a threshold voltage of a P-channel transistor. These values typically range from 1.0 V to 1.5 V in magnitude for this exemplary process technology. At time t 1  program data leads PGMDATA and PGMDATA_ of the memory cell column are driven high and low, respectively, to program a positive charge on floating gate  156 . At time t 2 , PGMDATA and PGMDATA_ return to 0 V. However, TG is pulled down to +V TP    832  by the P-channel transistor of the CMOS switch. Correspondingly, CG is pulled up to −V TN  by the N-channel transistor of the CMOS switch. Thus, TG and CG follow PGMDATA and PGMDATA_, respectively, but will only reach +V TP  or −V TN  depending on the previous voltage level of PGMDATA and PGMDATA_. 
     At time t 3  program data leads PGMDATA and PGMDATA_ of the memory cell column are driven low and high, respectively, to erase the positive charge on floating gate  156 . At time t 4 , PGMDATA and PGMDATA_ return to 0 V. However, TG is pulled up to −V TN    834  by the N-channel transistor of the CMOS switch. Correspondingly, CG is pulled down to +V TN    844  by the P-channel transistor of the CMOS switch. When any cell is on a selected row and PGMDATA and PGMDATA_ are 0 V, therefore, TG and CG will only reach +V TP  or −V TN  depending on the previous voltage level of PGMDATA and PGMDATA_. This produces a total cell stress equal to a sum of the magnitude of V TP +V TN  across the floating gate  156 . For normal operating parameters, this is approximately 2.5: V compared to a programming voltage of 10 V. At this level, there is negligible effect on the programmed or erased data state. Since ROWSEL and ROWSEL_ are both at 0 V, no more than 5 V appears across any transistor gate oxide of the CMOS switch. Furthermore, the gates of N-channel transistors  810  and  812  are at 0 V while the gates of N-channel transistors  811  and  813  are at −5 V. In this state, if V TG  is +5 V, transistor  810  acts as a voltage divider so that the common terminal between transistors  810  and  811  is −V TN . Likewise, if V CG  is +5 V, transistor  812  acts as a voltage divider so that the common terminal between transistors  812  and  813  is −V TN . Therefore, no more than 5 V appears across any transistor gate oxide of the equalization circuit. 
     Time t 5  and beyond represents a cell on an unselected row and a selected column. Here, EQ is high (+5 V), ROWSEL is low (−5 V), and ROWSEL_ is high (+5 V). Both CMOS switches are off. N-channel transistors  810 - 813  of the equalization circuit are on and drive TG and CG to ground. Thus, voltage levels of PGMDATA and PGMDATA_ have no effect on any memory cell in an unselected row. In this state, transistors  800  and  804  act as voltage dividers for either a positive or negative voltage of PGMDATA. Thus, common terminals between P-channel transistors  800  and  802  or between N-channel transistors  804  and  806  do not exceed a magnitude of V TN  or V TP . Therefore, no more than 5 V appears across any transistor gate oxide of the CMOS switch for any voltage level of PGMDATA and PGMDATA_. 
     Turning now to  FIG. 9A , there is a schematic diagram of a drive circuit that may be used with the memory cell of  FIG. 8A  to produce the equalization (EQ) signal. The circuit receives low voltage equalization signal (EQL), which operates between 0 V and +5 V, at the input terminal of inverter  900 . The output signal from inverter  900  at lead  922  is applied to the input terminal of inverter  902  as well as the source of P-channel transistor  904 . Inverter  902  produces an output signal at lead  924 , which is applied to the source of P-channel transistor  910 . Here, inverters  900  and  902  form a data input circuit. P-channel transistors  904  and  910  and N-channel transistors  906  and  912  form a voltage divider part of the drive circuit and have their control gates connected to reference or ground terminal  930 . Cross-coupled N-channel transistors  908  and  914  have current paths connected in series with N-channel transistors  906  and  912 , respectively. The control gate of N-channel transistor  908  is connected to the drain of N-channel transistor  914 . Correspondingly, the control gate of N-channel transistor  914  is connected to the drain of N-channel transistor  908 . A common source terminal of N-channel transistors  908  and  914  is connected to negative supply voltage terminal  920 . Capacitors  905  (C 1 ) and  907  (C 2 ) serve to couple a difference voltage from the output of inverters  900  and  902 , respectively, to the control gates of N-channel transistors  914  and  908 . These capacitors are preferably N-channel MOS transistors with common source/drain terminals. The equalization output signal (EQ) is taken from lead  926 . 
     In operation, a low input level at the input of inverter  900  produces a high level (+5 V) input signal at the source of P-channel transistor  904  and a low level input signal (0 V) at the source of P-channel transistor  910 . The control gate terminals of the voltage divider transistors ( 904 ,  906 ,  910 , and  912 ) are preferably grounded at a reference voltage of 0V. Therefore, the maximum steady state positive voltage at the control gate of either of N-channel transistors  908  and  914  is approximately an N-channel threshold voltage below ground (−Vtn). In this case, P-channel transistor  904  is on, and P-channel transistor  910  is off. Capacitors  905  and  907  respectively couple a difference voltage to the control gates of N-channel transistors  914  and  908 . These capacitors are optional but significantly increase the switching speed of the drive circuit. Cross coupled N-channel transistors  908  and  914  amplify the applied difference voltage and produce a low level (−5 V) equalization signal (EQ) at lead  926 . 
     A transition to a high level input signal at the input of inverter  900  produces a low level (0 V) input signal at the source of P-channel transistor  904  and a high level input signal (+5 V) at the source of P-channel transistor  910 . Thus, P-channel transistor  910  is on, and P-channel transistor  904  is off. Capacitors  905  and  907  again couple the applied difference voltage to the control gates of N-channel transistors  914  and  908 . Cross coupled N-channel transistors  908  and  914  amplify the applied difference voltage and produce a high level (+5 V) equalization signal (EQ) at lead  926 . Thus, the drive circuit switches between −5 V and +5 V in response to and input signal (EQL) transition between 0 V and  5  V. Advantageously, the maximum steady voltage across any gate oxide is substantially equal to 5 V plus an N-channel transistor threshold voltage (Vtn) or 60% to 65% of the program voltage (10 V). 
     Referring now to  FIG. 9B , there is a drive circuit that may be used with the memory cell of  FIG. 8A  to produce the program data (PGMDATA) and complementary program data (PGMDATA_) signals. The left part of the circuit is substantially the same as the previously described drive circuit of  FIG. 9A  except that the input is active low enable signal EN_. The right part of the circuit includes a data input part having a first inverter coupled to receive a low voltage program data signal (PGMDATAL) at lead  940  having a voltage range of preferably 0 V to +5 V. The first inverter produces a complementary signal on lead  942  that is applied to a second inverter to produce a second input signal on lead  944 . The first and second input signals are applied to tri state circuit  960 . Tri state circuit  960  relays a difference voltage from the data input part to amplifier part  962 . Together they produce program data signal PGMDATA and complementary program data signal PGMDATA_ on leads  982  and  980 , respectively. 
     In operation, when enable signal EN_ is high the circuit is in tri state mode and the signal on lead  946  is high (+5 V). This turns off both P-channel transistors and turns on both N-channel transistors having control gates connected to lead  946 . In addition, the signal on lead  950 , having a voltage range of −5 V to +5 V, is also high, thereby turning on both N-channel transistors having control gates connected to lead  950 . This couples leads  980  and  982  to ground and holds data signal PGMDATA and complementary program data signal PGMDATA_ at 0 V. Furthermore, when enable signal EN_ is high, the signal on lead  952  is −5 V and the signal on lead  954  is −Vtn or about −1 V to −1.5 V. This turns off both N-channel transistors having control gates connected to lead  952 . Correspondingly, both N-channel transistors having gates connected to lead  954  are on, thereby coupling the cross coupled N-channel transistor gates of the amplifier part  962  to −5 V. 
     When enable signal EN_ goes low, the drive circuit of  FIG. 9B  is enabled. Both P-channel transistors of the tri state part  960  are on and apply the difference voltage at leads  942  and  944  to leads  980  and  982 , respectively. Correspondingly, the signal on lead  952  is −Vtn and the signal on lead  954  is −5 V. Thus, the difference voltage at leads  980  and  982  is applied to the cross coupled N-channel transistor control gates of the amplifier part  962 . In this mode, amplifier part  962  operates as a low-going-high sense amplifier. Consequently, one of PGMDATA and PGMDATA_ is driven to +5 V and the other is driven to −5 V in response to the data state of input signal PGMDATAL. 
     A significant advantage of this drive circuit is that both the tri state part  960  and the amplifier part  962  employ N-channel voltage divider transistors. When the drive circuit is enabled, the N-channel transistors of tri state part  960  with control gates connected to lead  946  act as voltage dividers. Their source terminals are no more positive than −Vtn. Therefore, neither N-channel transistors having control gates connected to lead  946  nor to lead  950  have more than 5 V across their gate oxide in steady state when there are no related signal transitions. Likewise, the N-channel transistors of amplifier part  962  with control gates connected to lead  952  act as voltage dividers so that their source terminals are no more positive than −2Vtn. Therefore, none of the N-channel transistors in amplifier part  962  have more than 5 V plus Vtn across their gate oxide in steady state. This drive circuit advantageously produces sufficient programming voltage to operate the previously described memory cells but avoids significant stress on corresponding drive circuitry having the same gate oxide thickness. 
     Turning now to  FIG. 9C , there is another drive circuit that may be used with the memory cell of  FIG. 8A  to produce the program data (PGMDATA) and complementary program data (PGMDATA_) signals. The left part of the circuit, the data input part, and the tri state part  960  are substantially the same as the previously described drive circuit of  FIG. 9B . Here, however, the control gates of N-channel voltage divider transistors of amplifier part  964  are grounded so their sources are no more positive than one N-channel transistor threshold below ground potential (−Vtn). Amplifier part  964  also includes N-channel equalization transistor  956  having a control gate connected to lead  950  and N-channel activation transistor  958  having a control gate connected to lead  952 . 
     Operation of the drive circuit when enable signal EN_ is high is substantially the same as previously described with regard to  FIG. 9B . Here, however, the signal on lead  950  is high (+5 V) so N-channel transistor  956  is on and equalizes the voltage of the cross coupled N-channel transistor control gates to near −Vtn. The signal on lead  952  is −5 V so N-channel activation transistor  958  is off. 
     When enable signal EN_ goes low, the drive circuit of  FIG. 9C  is enabled. The tri state part  960  applies the difference voltage at leads  942  and  944  to leads  980  and  982 , respectively. Correspondingly, the signal on lead  950  goes to −5 V and turns off N-channel equalization transistor  956 . The −Vtn signal on lead  952  turns on N-channel activation transistor  958 . This amplifies the difference voltage at leads  980  and  982  and operates as a high-going-low sense amplifier. Consequently, one of PGMDATA and PGMDATA_ is driven to +5 V and the other is driven to −5 V in response to the data state of input signal PGMDATAL. 
     This drive circuit offers substantially the same advantages as previously described with regard to  FIG. 9B . When the drive circuit is enabled, the N-channel transistors of amplifier part  964  with control gates connected to ground act as voltage dividers so that their source terminals are no more positive than −Vtn. Therefore, none of the N-channel transistors in amplifier part  964  have more than 5 V plus Vtn across their gate oxide in steady state. This drive circuit advantageously produces sufficient programming voltage to operate the previously described memory cells but avoids stress on corresponding drive circuitry having the same gate oxide thickness. 
     Still further, while numerous examples have thus been provided, one skilled in the art should recognize that various modifications, substitutions, or alterations may be made to the described embodiments while still falling with the inventive scope as defined by the following claims. For example, inventive concepts of the present invention are readily adapted to alternative designs and voltage levels as would be apparent to one of ordinary skill in the art having access to the instant specification. For example, previously described drive circuits might be redesigned to employ cross coupled P-channel transistors and P-channel voltage divider transistors rather than cross coupled N-channel transistors and N-channel voltage divider transistors. Likewise, BiCMOS processes might mix bipolar and MOS transistors to produce the previously described drive circuits. Additionally, programming voltages might range from 0 V to 10 V or from 0 V to −10V rather than from −5V to 5 V. Other combinations will be readily apparent to one of ordinary skill in the art having access to the instant specification.