Abstract:
A method and apparatus for performing chip level equalization (CLE) using joint processing to enhance performance and system throughput using a transmitter having a plurality of transmit antennas and a receiver having a plurality of receive antennas. A channel response matrix is formed between the transmit antennas and the receive antennas to generate a joint channel correlation matrix between the transmit antennas and the receive antennas using a block-FFT (B-FFT) decomposition of the channel response matrix. Estimates of transmitted chip sequences from each of the transmit antennas are generated using minimum mean square error (MMSE) and the joint channel correlation matrix are combined. The combined estimate of the transmitted chip sequences are despread to recover transmitted data.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS  
       [0001]     This application claims the benefit of U.S. Provisional Patent Application Nos. 60/636,345 filed Dec. 14, 2004 and 60/642,383 filed Jan. 7, 2005, which are incorporated by reference as if fully set forth. 
     
    
     FIELD OF INVENTION  
       [0002]     The present invention is related to a wireless communication receiver. More particularly, the present invention relates to a receiver that processes space-time transmit diversity (STTD), closed loop transmit diversity for transmit adaptive antennas and receiver diversity with over-sampling and fast Fourier transform (FFT)-based chip level equalization (CLE) using joint processing.  
       BACKGROUND  
       [0003]     CLE is a candidate for use in advanced receivers in wireless communication systems for high data rate services such as high speed downlink packet access (HSDPA). CLE-based receivers, such as those used in wireless transmit/receive units (WTRUs), are used more often than Rake receivers in advanced receivers due to their superior performance.  
         [0004]     Receive diversity using two or more receive antennas provides high performance by improving the reception quality of signals. Over-sampling is also used to improve the reception performance by rectifying performance degradations caused by timing errors or sampling errors. In addition, transmit adaptive antennas are used to improve signal degradations caused by fading, and thus improve data detection performance at the receiver and enhance the system throughput.  
         [0005]     In conventional receivers which implement equalization, each channel that corresponds to an antenna is equalized independently of other channels that correspond to other antennas. However, these type of receivers usually experience significant performance degradations due to mutual channel interference from one antenna to another that cannot be eliminated or cancelled. Therefore, there is a need for receivers which implement CLE such that mutual channel interference is reduced or eliminated.  
       SUMMARY  
       [0006]     The present invention is related to a method and apparatus for performing CLE using joint processing to enhance performance and system throughput using a transmitter having a plurality of transmit antennas and a receiver having a plurality of receive antennas. A channel response matrix is formed between the transmit antennas and the receive antennas to generate a joint channel correlation matrix between the transmit and the receive antennas using a block-FFT (B-FFT) decomposition of the channel response matrix. Estimates of transmitted chip sequences from each of the transmit antennas are generated using minimum mean square error (MMSE) and the joint channel correlation matrix are combined. The combined estimate of the transmitted chip sequences is despread to recover transmitted data. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0007]     A more detailed understanding of the invention may be had from the following description of a preferred embodiment, given by way of example and to be understood in conjunction with the accompanying drawings wherein:  
         [0008]      FIG. 1  is a block diagram of a transmitter for supporting closed loop mode transmit diversity for dedicated physical channel (DPCH) transmission in accordance with the present invention;  
         [0009]      FIGS. 2A and 2B , taken together, are an exemplary block diagram of a receiver implementing B-FFT-based CLE using joint processing with transmit and receive diversity at twice the chip rate in accordance with the present invention;  
         [0010]      FIG. 3  shows a space time transmit diversity (STTD) encoder for quadrature phase shift keying (QPSK);  
         [0011]      FIG. 4  shows an STTD encoder for 16 quadrature amplitude modulation (16 QAM); and  
         [0012]      FIGS. 5A and 5B , taken together, are an exemplary block diagram of a receiver implementing B-FFT-based CLE using joint processing in STTD and receive diversity with over sampling in accordance with the present invention. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0013]     The present invention will be described with reference to the drawing figures wherein like numerals represent like elements throughout.  
         [0014]     Hereafter, the terminology “WTRU” includes but is not limited to a user equipment (UE), a mobile station, a fixed or mobile subscriber unit, a pager, or any other type of device capable of operating in a wireless environment.  
         [0015]     The features of the present invention may be incorporated into an integrated circuit (IC) or be configured in a circuit comprising a multitude of interconnecting components.  
         [0016]     The present invention provides a method and apparatus for implementing an advanced wireless receiver using CLE and joint processing. The joint processing eliminates or reduces mutual channel interference and enhances data detection performance and system throughput. The joint processing-based CLE in accordance with the present invention utilizes transmit diversity and receive diversity with over-sampling. The over-sampling is preferably at twice the chip rate, but the sampling rate may be at any rate. Compared with the receiver using individual equalizers, where each equalizer is dedicated for one antenna, the joint processing-based CLE considers the mutual interference between antennas and eliminates the mutual interferences using joint approaches. Furthermore, the joint processing-based CLE in accordance with the present invention uses B-FFT techniques to realize efficient implementation. The B-FFT and joint processing-based CLE in the present invention has the same number of FFT operations as compared to a prior art receiver without joint processing.  
         [0017]      FIG. 1  is a block diagram of a transmitter  100  for supporting closed loop mode transmit diversity for dedicated physical channel (DPCH) transmission in accordance with the present invention. In a closed loop mode transmit diversity, a WTRU sends a feedback signaling message (FSM) to the UMTS terrestrial radio access network (UTRAN) to maximize the received power of the WTRU. Two different closed loop modes, (closed loop modes  1  and  2 ) are defined. The use of the two closed loop modes is controlled via higher layer signaling.  
         [0018]     As shown in  FIG. 1 , a DPCH data sequence  102 , (including a dedicated physical control channel (DPCCH) data sequence and a dedicated physical data channel (DPDCH) data sequence is despread and descrambled by multiplying the DPCH data sequence  102  with a spreading code and scrambling code  104  via a multiplier  106  to generate a spread complex valued signal  108 . The spread complex valued signal  108  is fed into multipliers  110 ,  112 , each of which multiplies the spread complex valued signal  108  by a first antenna specific weight factor  114 , w 1 , and a second antenna specific weight factor  116 , w 2 , respectively. The weight factors  114 ,  116  are complex valued signals, (i.e., w i =a i +jb i ), which are generated by a weight generator  118  based on a feedback information (FBI) message  120  from an uplink DPCCH.  
         [0019]     As shown in  FIG. 1 , the resulting signals  122 ,  124  output from the multipliers  110 ,  112  are respectively summed with respective common pilot channels (CPICHs)  126 ,  128  via a respective summer  130 ,  132  to generate transmission signals  134 ,  136  which are transmitted by respective antennas  138 ,  140 .  
         [0020]     The weight factors  114 ,  116  correspond to phase adjustments in a closed loop mode  1  and phase/amplitude adjustments in a closed loop mode  2 . For the closed loop mode  1 , different, (preferably orthogonal), dedicated pilot symbols in the DPCCH are transmitted by the antennas  138 ,  140 . For the closed loop mode  2 , the same dedicated pilot symbols in the DPCCH are transmitted by the antennas  138 ,  140 .  
         [0021]     The transmitter  100  uses the CPICH signals  126 ,  128  transmitted from the antenna  138  and the antenna  140  to calculate the phase adjustment to be applied at the UTRAN to maximize the received power of a WTRU including the receiver  200  of  FIGS. 2A and 2B . In each time slot, the receiver  200  calculates the optimum phase adjustment, φ, for antenna  140 , which is then quantized into φ Q  having two possible values as follows:  
               ϕ   Q     =     {             π   ,             if   ⁢           ⁢     π   /   2       &lt;     ϕ   -       ϕ   r     ⁡     (   i   )         ≤     3   ⁢     π   /   2                   0   ,         otherwise         ;     
     ⁢   where               Equation   ⁢           ⁢     (   1   )                     ϕ   r     ⁡     (   i   )       =     {           0   ,             i   =   0     ,   2   ,   4   ,   6   ,   8   ,   10   ,   12   ,   14                 π   /   2     ,             i   =   1     ,   3   ,   5   ,   7   ,   9   ,   11   ,   13                     Equation   ⁢           ⁢     (   2   )               
 
 If φ Q =0, a command ‘0’ is sent to the UTRAN using the FSM ph  field and if φ Q =π, a command ‘1’ is sent to the UTRAN using the FSM ph  field. 
 
         [0022]     Due to rotation of the constellation at the WTRU in the closed loop mode  1 , the UTRAN interprets the received commands according to table  1  which shows the mapping between phase adjustment, φ i , and the received feedback command for each uplink slot.  
                                                                                                                                                               TABLE 1                                       Slot #                0   1   2   3   4   5   6   7   8   9   10   11   12   13   14                        FSM   0   0    π/2   0    π/2   0    π/2   0    π/2   0    π/2   0    π/2   0    π/2   0           1   π   −π/2   π   −π/2   π   −π/2   π   −π/2   π   −π/2   π   −π/2   π   −π/2   π                  
 
         [0023]     The weight  116 , w 2 , is then calculated by averaging the received phases over  2  consecutive slots as follows:  
                 w   2     =           ∑     i   =     n   -   1       n     ⁢     cos   ⁡     (     ϕ   i     )         2     +     j   ⁢         ∑     i   =     n   -   1       n     ⁢     sin   ⁡     (     ϕ   i     )         2           ;           Equation   ⁢           ⁢     (   3   )               
 
 where φ i ε0,π,π/2,−π/2}. For antenna  1 , w 1  is constant w 1 =1/√{square root over (2)}. 
 
         [0024]     The phase and amplitude are both adjusted in the closed loop mode  2 . The adjustments are based on the commands received in the FSM and are summarized in Tables 2 and 3 for the power and phase adjustments respectively.  
                       TABLE 2                       FSM po     Power_ant1   Power_ant2                   0   0.2   0.8       1   0.8   0.2                  
 
         [0025]    
       
         
               
               
               
             
           
               
                   
                 TABLE 3 
               
               
                   
                   
               
               
                   
                   
               
               
                   
                 FSM ph   
                 Phase difference between antennas (radians) 
               
               
                   
                   
               
             
             
               
                   
                 000 
                 π 
               
               
                   
                 001 
                  −3π/4       
               
               
                   
                 011 
                 −π/2   
               
               
                   
                 010 
                 −π/4   
               
               
                   
                 110 
                 0 
               
               
                   
                 111 
                 π/4 
               
               
                   
                 101 
                 π/2 
               
               
                   
                 100 
                 3π/4  
               
               
                   
                   
               
             
          
         
       
     
         [0026]     Antenna  138  transmits data symbols using weight coefficient w 1   (k)    112  and antenna  140  transmits data symbols using weight coefficient w 2   (k)    116  for the k th  channelization code.  
         [0027]     The received signal can be expressed as follows:  
                 r   _     =         H   1     ⁡     (       ∑     k   =   1     K     ⁢       w   1     (   k   )       ⁢       s   _     k         )       +       H   2     ⁡     (       ∑     k   =   1     K     ⁢       w   2     (   k   )       ⁢       s   _     k         )       +   n       ;           Equation   ⁢           ⁢     (   4   )               
 
 where H 1  and H 2  is the channel response matrix corresponding to the first and second (diversity) transmit antennas, respectively. The transmitted chip sequences are related by the spreading code matrix C as s k =C k {right arrow over (d)} k  for the k th  code. The weighted composite chip sequences are  
           t   →     1     =         ∑     k   =   1     K     ⁢       w   1     (   k   )       ⁢       s   _     k     ⁢           ⁢   and   ⁢           ⁢       t   →     2         =       ∑     k   =   1     K     ⁢       w   2     (   k   )       ⁢         s   _     k     .               
 
 Equation (4) can be rewritten as follows: 
 
   r =H   1   {right arrow over (t)}   1   +H   2   {right arrow over (t)}   2   + n .    Equation (5) 
 
         [0028]     The weighted composite chip sequences {right arrow over (t)} 1  and {right arrow over (t)} 2  can be demodulated using MMSE solution such that: 
 
   {circumflex over (t)}   =( H   H   H+σ   2   I ) −1   H   H     r .    Equation (6) 
 
 The vector {right arrow over ({circumflex over (t)})} is the estimated composite chip sequences and is expressed by {right arrow over ({circumflex over (t)})}=[{right arrow over (t)} 1 ,{right arrow over (t)} 2 ] T . 
 
         [0029]     In the presence of receive diversity and over-sampling, the channel response matrix H can be written as follows:  
               H   =     [           H     1   ,   o                 H     1   ,   e                 H     2   ,   o                 H     2   ,   e             ]       ;           Equation   ⁢           ⁢     (   7   )               
 
 where H i,o  and H i,e  i=1, . . . N are the channel response matrix of the i th  receiving antenna for odd and even sample sequences, respectively. Typically, N=2 for receive diversity and twice the chip rate sampling is used. However, N can be any number and the sampling rate can be any rate. For simplicity and illustration purposes, the present invention will be explained with reference to N=2 and twice the chip rate sampling hereinafter. In the presence of transmit adaptive antennas and receive diversity (N=2) with twice the chip rate over-sampling, the channel response matrix H can be written as follows:  
               H   =     [           H     1   ,   o       (   1   )             H     1   ,   o       (   2   )                 H     1   ,   e       (   1   )             H     1   ,   e       (   2   )                 H     2   ,   o       (   1   )             H     2   ,   o       (   2   )                 H     2   ,   e       (   1   )             H     2   ,   e       (   2   )             ]       ;           Equation   ⁢           ⁢     (   8   )               
 
 where H i,o   (j)  and H i,e   (j)  are the channel response matrix of the i th  receive antenna and the j th  transmit antenna for odd and even sample sequences, respectively. 
 
         [0030]     The estimated data symbols {right arrow over (d)} 1  and {right arrow over (d)} 2  can be simply obtained by multiplying the equalized composite chip sequences with complex conjugate of weights for both antennas, adding them up and despreading the added results as follows: 
 
 d   k   =C   k   H ( w   1   (k)   *{right arrow over (t)}   1   +w   2   (k) *{right arrow over (t)} 2 ).   Equation (9) 
 
         [0031]     B-FFT is used to realize the joint processing. H i,o  represents the channel response matrix for the i th  receive antenna and odd sample sequences and for both transmit antenna. H i,o  can be expressed as follows: 
 
 H   i,o   =[H   1,o   (1)  H i,o   (2) ].   Equation (10) 
 
         [0032]     The channel response matrix H i,o  can be further expressed in details in terms of channel coefficients as follows:  
               H     1   ,   o       =       [           h     1   ,   0             h     2   ,   0                                                                                                     h     1   ,   1             h     2   ,   1             h     1   ,   0             h     2   ,   0                                                                           ⋮       ⋮         h     1   ,   1             h     2   ,   1                                                                           ⋮       ⋮       ⋮       ⋮                                                                         h     1   ,     W   -   2               h     2   ,     W   -   2             ⋮       ⋮       ⋰                                                             h     1   ,     W   -   1               h     2   ,     W   -   1               h     1   ,     W   -   2               h     1   ,     W   -   2                         ⋰                                                                         h     1   ,     W   -   1               h     1   ,     W   -   1                                     ⋰                                   ⋮       ⋮                                                                                                           ⋮                                                                     h     1   ,   0             h     2   ,   0                                                               ⋰                                 h     1   ,   1             h     2   ,   1                                                                           ⋰                   ⋮       ⋮                                                                                   ⋰       ⋮       ⋮           ⋮       ⋮                                                                     h     1   ,     W   -   2               h     2   ,     W   -   2                                                                                                       h     1   ,     W   -   1               h     2   ,     W   -   1               ]     .             Equation   ⁢           ⁢     (   11   )               
 
 H 1,o  is expressed by the channel coefficients with pre-ordering of columns of channel matrix to transform the original matrix into a block circular matrix for channel response matrix H and to enable efficient B-FFT computations. Similarly, H 2,o , H 1,e  and H 2,e  can be expressed in the same form that enables the B-FFT. 
 
         [0033]     Each block is defined as follows: H i =[h 1,i  h 2,i ], i==0,1,2, . . . ,W−1. H 1,o  can then be expressed as follows:  
                 H     1   ,   o       =     [           H   0                                                   H   1           H   0                                     ⋮         H   1                                       H     W   -   1           ⋮                                                 H     W   -   1                                                               ⋰                                                             H   0                                                   H   1                                                 ⋮                                                 H     W   -   1             ]       ;           Equation   ⁢           ⁢     (   12   )               
 
 Where each H i  is a matrix of size one by two. 
 
         [0034]     F (P)  and F (K)  are B-FFT matrices of size P×P and K×K, respectively. The matrix H 1,o  can be decomposed by B-FFT in an extended manner as follows: 
 
 H   1,o   =F   (P)   −1   Λ   H   F   (K)    Equation (13) 
 
 where 
 
 F   (P)   =F   L   I   P ;   Equation (14) 
 
 and 
 
 F   (K)   =F   L           I K ; Equation   (15) 
 
 where F L  is the L-point FFT matrix, I P  and I K  are the identity matrix of size P and K, respectively, and          is a kronecker product. For example, L=256 or 512, P=1 and K=2. It should be noted that the foregoing numbers are provided as an example and any other numbers may be implemented. L is scalable for more efficient implementation. Λ H  is a block-diagonal matrix whose diagonal blocks are F (K) H(:,1: K). 
 
Λ H =diag( F   (K)   H (:,1 : K )).   Equation (16) 
 
Also H 1,o   H   =F   (K)   −1 Λ* H   F   (P) ;   Equation (17) 
 
 H   1,o   H   H   1,o   =F   (K)   −1 Λ* H Λ H   F   (K) ;   Equation (18) 
 
and  H   1,o   H   {right arrow over (r)}   1,o   =F   (K)   −1 Λ* H   F   (P)   {right arrow over (r)}   1,o .   Equation (19) 
 
         [0035]     The transmitted data sequence s can be solved by the following equations: 
 
 y =Λ H   H   F   (P)   {right arrow over (r)};    Equation (20) 
 
 y=κ   H   H κ H     x ; and    Equation (21) 
 
 t=F   (K)   −1     x .    Equation (22) 
 
         [0036]     In general, x can be solved block by block using Cholesky decomposition. Since the block size is very small (only 2×2) here in consideration, a direct matrix inverse of each block can be performed without using Cholesky decomposition. A similar approach can also be developed using time domain channel correlation matrix R=H H H.  
         [0037]     The correlation matrix R can be decomposed by B-FFT as follows: 
 
 R=F   (P)   −1 Λ H   F   (K) ;   Equation (23) 
 
 where Λ R  is a block-diagonal matrix whose diagonal blocks are F (K) R(:,1: K) 
 
         [0038]     The above procedure is performed for H 1,o , H 2,o , H 1,e  and H 2,e  to develop the entire solution of joint processing and B-FFT is used to realize the joint processing for transmit adaptive antenna and receive diversity.  
         [0039]     The detected data symbols of two transmit data sequences using joint processing are as follows:  
               t   →     =         [         ∑     i   =   1     N     ⁢     (         H     i   ,   o     H     ⁢     H     i   ,   o         +       H     i   ,   e     H     ⁢     H     i   ,   e           )       +       σ   2     ⁢   I       ]       -   1       ·       [         ∑     i   =   1     N     ⁢       H     i   ,   o     H     ⁢       r   →       i   ,   o           +       H     i   ,   e     H     ⁢       r   →       i   ,   e           ]     .               Equation   ⁢           ⁢     (   24   )               
 
         [0040]     The realization of joint processing using B-FFT are as follows:  
               t   →     =           F     (   K   )       -   1       ⁡     [         ∑     i   =   1     N     ⁢     (         Λ     i   ,   o     *     ⁢     Λ     i   ,   o         +       Λ     i   ,   e     *     ⁢     Λ     i   ,   e           )       +       σ   2     ⁢   I       ]         -   1       ·             [         ∑     i   =   1     N     ⁢       Λ     i   ,   o     *     ⁢     F     (   P   )       ⁢       r   →       i   ,   o           +       Λ     i   ,   e     *     ⁢     F     (   P   )       ⁢       r   →       i   ,   e           ]     .                 Equation   ⁢           ⁢     (   25   )               
 
         [0041]     By letting T and {right arrow over (y)} represent as follows:  
                 T   =         ∑     i   =   1     N     ⁢     (         Λ     i   ,   o     *     ⁢     Λ     i   ,   o         +       Λ     i   ,   e     *     ⁢     Λ     i   ,   e           )       +       σ   2     ⁢   I         ;     ⁢     
     ⁢     and   ,             Equation   ⁢           ⁢     (   26   )                     y   →     =         ∑     i   =   1     N     ⁢       Λ     i   ,   o     *     ⁢     F     (   P   )       ⁢       r   →       i   ,   o           +       Λ     i   ,   e     *     ⁢     F     (   P   )       ⁢       r   →       i   ,   e             ;           Equation   ⁢           ⁢     (   27   )               
 
 Equation (25) can be rewritten as follows: 
 
 T·F   (K)   {right arrow over (t)}={right arrow over (y)}.    Equation (28) 
 
         [0042]     {right arrow over (x)}=F (K) {right arrow over (t)} by Equation (22). Therefore, Equation (27) can be rewritten as follows: 
 
T{right arrow over (x)}={right arrow over (y)}.   Equation (29) 
 
         [0043]     The unknown {right arrow over (x)} is solved first. Once {right arrow over (x)} are solved, inverse FFT is performed on {right arrow over (x)} to obtain the composite chip sequences to be estimated as follows: 
 
 {right arrow over (t)}=F   (K)   −1   {right arrow over (x)}.    Equation (30) 
 
         [0044]     F (K)   −1  is exchangeable with F (K)  as follows:  
               F     (   k   )       -   1       =       1   L     ⁢       F     (   K   )     *     .               Equation   ⁢           ⁢     (   31   )               
 
         [0045]      FIGS. 2A and 2B , taken together, are an exemplary block diagram of a receiver  200  implementing B-FFT-based CLE using joint processing with transmit and receive diversity with two transmit antennas and two receive antennas at twice the chip rate in accordance with the present invention. As explained hereinbefore, any number of transmit and receive antennas and any sampling rate may be used. In this example, for a received signal r, four sample streams  202   1 - 202   4  are generated from two receive antennas (not shown). From the sample streams  202   1 - 202   4 , channel responses between a first transmit antenna and two receive antennas for even and odd sample sequences {right arrow over (h)} (1)    206   1 - 206   4  and channel responses between a second transmit antenna and two receive antennas for even and odd sample sequences {right arrow over (h)} (2)    206   5 - 206   8  are generated by a channel estimator (not shown).  
         [0046]     The sample streams  202   1 - 202   4  are processed by FFT units  204   1 - 204   4  to be converted into frequency domain data, respectively. The channel response vectors  206   1 - 206   8  are processed by FFT units  208   1 - 208   8 , respectively to generate frequency domain channel response vectors  210   1 - 210   8 . Complex conjugates  214   1 - 214   8  of the frequency domain channel response vectors  210   1 - 210   8  are generated by complex conjugate units  212   1 - 212   8 , respectively. The frequency domain sample streams  216   1 - 216   4  and complex conjugates  214   1 - 214   8  of the frequency domain channel response vectors  210   1 - 210   8  are multiplied by element-wise multipliers  218   1 - 218   8 , respectively. The multiplication results for the first transmit antenna  220   1 - 220   4  are combined by a combiner  222   1  and the multiplication results for the second transmit antenna  220   5 - 220   8  are combined by a combiner  222   2 . The combined results y (1) , y (2) , ( 224   1 ,  224   2 ), correspond to the output of Equation (20) (or Equation (27)).  
         [0047]     The frequency domain channel response vectors  210   1 - 210   8  and a noise variance value  232  enter a joint channel correlation generator  230 . Equation (18) depicts the function of generator  230  for channel correlation generation that occurs in frequency domain. The function of processor  240  is depicted by Equations (18), (20), (21) and (22) for solving the linear systems. The joint channel correlation generator  230  generates joint channel correlation matrix  234   1 - 234   4  between two transmit antennas and two receive antennas and even and odd sample stream. The joint channel correlation matrixes  234   1 - 234   4  are combined by a combiner  236  and the combined joint channel correlation matrix  238 , which corresponds to T in Equation (26), enters a processor  240 .  
         [0048]     The processor  240  receives as an input the combined joint channel correlation matrix  238  and two combined results y (1) , y (2)    224   1 ,  224   2  and generates estimates of the transmitted chip sequences by solving the 2×2 linear systems of Equation (29). The estimates of the transmitted chip sequences  242   1 ,  242   2  undergo transmit adaptive antenna processing by being multiplied with complex conjugates  248   1 ,  248   2  of weight factors  244   1 ,  244   2  generated by complex conjugate units  246   1 ,  246   2 , by element-wise multipliers  218   9  and  218   10 , respectively. The two multiplier outputs  250   1 ,  250   2  are soft combined by a summer  252  and the combined output  254  is processed by an IFFT unit  256  to be converted into time domain signals  258 . Then, the time domain signals  258  are processed by a despreader  260  to generate a data symbol estimate  262 .  
         [0049]     The present invention may be implemented with STTD. For STTD, a first antenna transmits {right arrow over (d)} 1  and a second antenna transmits {right arrow over (d)} 2 , where {right arrow over (d)} 1  and {right arrow over (d)} 2  are STTD encoded data sequences.  FIG. 3  shows the STTD encoded data sequences for QPSK, such that {right arrow over (d)} 1 =[b 0  b 1  b 2  b 3 ] T  and {right arrow over (d)} 2 =[{overscore (b)} 2  b 3  b 0  {overscore (b)} 1 ] T .  FIG. 4  shows the STTD encoded data sequences for 16 QAM such that {right arrow over (d)} 1 =[b 0  b 1  b 2  b 3  b 4  b 5  b 6  b 7 ] T  and {right arrow over (d)} 2 =[{overscore (b)} 4  b 5  b 6  b 7  b 0 {overscore (b)} 1  b 2  b 3 ] T .  
         [0050]     The received signal at the receiver can be expressed as follows: 
 
   r =H   1   s   1   +H   2   s   2   + n ;    Equation (32) 
 
 where H 1  and H 2  is the channel response matrix corresponding to the first and second diversity antennas, respectively. The chip and STTD encoded symbol sequences are related by the spreading code matrix C as s 1 =C{right arrow over (d)} 1  and s 2 =C{right arrow over (d)} 2 . 
 
         [0051]     The chip sequences s 1  and s 2  can be demodulated at the receiver using MMSE such that: 
 
 ŝ =( H   H   H+σ   2   I ) −1   H   H     r .    Equation (33) 
 
         [0052]     In the presence of receive diversity and over-sampling, the channel response matrix H can be expressed by Equation (7) and in the presence of STTD transmit diversity and receive diversity (N=2) with twice the chip rate over-sampling, the channel response matrix H can be expressed by Equation (8).  
         [0053]     The STTD encoded data symbols {right arrow over (d)} 1  and {right arrow over (d)} 2  can be simply obtained by de-spreading the equalized chip sequences. Because data sequences b i , i=0,1,2, . . . ,7 are detected in both STTD encoded data vectors {right arrow over (d)} 1  and {right arrow over (d)} 2 , the STTD decoding and soft combining are used to achieve diversity gain and improve performance such as: 
 
 d=α   1 ·sign( b   i,ant1 )+α 2 ·sign( b   i,ant2 );   Equation (34) 
 
 where the notation sign ( ) represents the sign changes according to STTD decoding rules and modulation types, such as QPSK and 16 QAM. 
 
         [0054]     For QPSK, the STTD decoding are described as follows:  
         [0055]     Antenna  1 : 
 
sign( b   i,ant1 )= b   i,ant1 , for all  i  
 
         [0056]     Antenna  2 : 
 
sign( b   i,ant2 )= b   i,ant2 , if  i= 0,3 
 
sign( b   i,ant2 )=− b   i,ant2 , else (or  i= 1,2) 
 
         [0057]     For 16 QAM, the STTD decoding are as follows:  
         [0058]     Antenna  1 : 
 
sign( b   i,ant1 )= b   i,ant1 , for all  i  
 
         [0059]     Antenna  2 : 
 
sign( b   i,ant1 )= b   i,ant2 , if  i= 0,2,3,5,6,7 
 
sign( b   i,ant2 )=− b   i,ant2 , else (or  i= 1,4) 
 
         [0060]     For equal gain soft combining, the weight coefficients are α=α 2 =1. For maximal ratio combining (MRC), the weight coefficients α n , n=1,2 are preferably as follows:  
                 α   n     =           ∑   i     ⁢            h     n   ,   i            2             ∑   i     ⁢            h     1   ,   i            2       +       ∑   i     ⁢          h     2   ,   i                      ,     n   =   1     ,   2.           Equation   ⁢           ⁢     (   35   )               
 
         [0061]     B-FFT is used to realize the joint processing. For example, H i,o  representing the channel response matrix for the i th  receive antenna and odd sampled sequences and for both transmit antenna can be expressed as follows: 
 
 H   i,o   =[H   i,o   (1)  H i,o   (2) ].   Equation (36) 
 
         [0062]     The channel response matrix H i,o  can be expressed in details by Equation (11) in terms of channel coefficients and can also be expressed by Equation (12). The matrix H i,o  can be decomposed by B-FFT by Equations (13)-(15).  
         [0063]     The transmitted data sequence s can be estimated by the following equations: 
 
   y =F   (P)     r ;    Equation (37) 
 
Λ H   H     y =Λ   H   H Λ H     x ;    Equation (38) 
 
 s=F   (K)   −1     x .    Equation (39) 
 
         [0064]     In general, the x can be solved block by block using Cholesky decomposition. Since the block size is very small (only 2×2) for the example under consideration, a solution using a direct matrix inverse of each block can be performed without using Cholesky decomposition. A similar approach can also be developed using a time domain channel correlation matrix R=H H H. Same procedure is repeated for H 1,o , H 2,o , H 1,e  and H 2,e  to develop the entire solution of joint processing and B-FFT is used to realize the joint processing for STTD and receive diversity.  
         [0065]     The detected data symbols of two transmit data sequences using joint processing are expressed as follows:  
                 d   -&gt;     Tx     =         [         ∑     i   =   1     N     ⁢     (         H     i   ,   o     H     ⁢     H     i   ,   o         +       H     i   ,   e     H     ⁢     H     i   ,   e           )       +       σ   2     ⁢   I       ]       -   1       ·             [         ∑     i   =   1     N     ⁢       H     i   ,   o     H     ⁢       r   -&gt;       i   ,   o           +       H     i   ,   e     H     ⁢       r   -&gt;       i   ,   e           ]     .                 Equation   ⁢           ⁢     (   40   )               
 
         [0066]     The realization of joint processing using B-FFT are as follows:  
                 d   -&gt;     Tx     =           F     (   K   )       -   1       ⁡     [         ∑     i   =   1     N     ⁢     (         Λ     i   ,   o     *     ⁢     Λ     i   ,   o         +       Λ     i   ,   e     *     ⁢     Λ     i   ,   e           )       +       σ   2     ⁢   I       ]         -   1       ·             [         ∑     i   =   1     N     ⁢       Λ     i   ,   o     *     ⁢     F     (   P   )       ⁢       r   →       i   ,   o           +       Λ     i   ,   e     *     ⁢     F     (   P   )       ⁢       r   →       i   ,   e           ]     .                 Equation   ⁢           ⁢     (   41   )               
 
         [0067]     By letting R fft  and {right arrow over (y)} represent as follows:  
                   R   fft     =         ∑     i   =   1     N     ⁢     (         Λ     i   ,   o     *     ⁢     Λ     i   ,   o         +       Λ     i   ,   e     *     ⁢     Λ     i   ,   e           )       +       σ   2     ⁢   I         ;     ⁢     
     ⁢   and           Equation   ⁢           ⁢     (   42   )                     y   →     =         ∑     i   =   1     N     ⁢       Λ     i   ,   o     *     ⁢     F     (   P   )       ⁢       r   →       i   ,   o           +       Λ     i   ,   e     *     ⁢     F     (   P   )       ⁢       r   →       i   ,   e             ,           Equation   ⁢           ⁢     (   43   )               
 
 the Equation (41) can be rewritten as follows: 
 
 R   fft   F   (K)   {right arrow over (d)}   Tx   ={right arrow over (y)}.    Equation (44) 
 
         [0068]     Furthermore, by letting {right arrow over (x)}=F (K) {right arrow over (d)} Tx , a linear system is obtained such that: 
 
R fft   {right arrow over (x)}={right arrow over (y)}.    Equation (45) 
 
         [0069]     After solving the unknown {right arrow over (x)}, an inverse FFT is performed on {right arrow over (x)} to obtained the data symbols to be estimated as follows: 
 
 {right arrow over (d)}   Tx   =F   (K)   −1   {right arrow over (x)}.    Equation (46) 
 
         [0070]     F (K)   −1  is exchangeable with F (K)  as follows:  
               F     (   K   )       -   1       =       1   L     ⁢       F     (   K   )     *     .               Equation   ⁢           ⁢     (   47   )               
 
         [0071]      FIGS. 5A and 5B , taken together, are a block diagram of a receiver  300  implementing B-FFT-based CLE using joint processing in STTD and receive diversity with over sampling in accordance with the present invention. As explained hereinbefore, any number of transmit and receive antennas and any sampling rate may be used. In this example, for a received signal r, four sample streams  302   1 - 302   4  are generated from two receive antennas (not shown). From the sample streams  302   1 - 302   4 , channel responses between a first transmit antenna and two receive antennas for even and odd sample sequences {right arrow over (h)} (1)    306   1 - 306   4  and channel responses between a second transmit antenna and two receive antennas for even and odd sample sequences {right arrow over (h)} (2)    306   5 - 306   8  are generated by a channel estimator (not shown).  
         [0072]     The sample streams  302   1 - 302   4  are processed by FFT units  304   1 - 304   4  to be converted into frequency domain data, respectively. The channel response vectors  306   1 - 306   8  are processed by FFT units  308   1 - 308   8 , respectively to generate frequency domain channel response vectors  310   1 - 310   8 . Complex conjugates  314   1 - 314   8  of the frequency domain channel response vectors  310   1 - 310   8  are generated by complex conjugate units  312   1 - 312   8 , respectively. The frequency domain sample streams  316   1 - 316   4  and complex conjugates  314   1 - 314   8  of the frequency domain channel response vectors  310   1 - 310   8  are multiplied by element-wise multipliers  318   1 - 318   8 , respectively. The multiplication results for the first transmit antenna  320   1 - 320   4  are combined by a combiner  322   1  and the multiplication results for the second transmit antenna  320   5 - 320   8  are combined by a combiner  322   2 . The combined results y (1) , y (2) , ( 324   1 ,  324   2 ), which correspond to the output of Equation (48).  
         [0073]     The frequency domain channel response vectors  310   1 - 310   8  and a noise variance value  332  enter a joint channel correlation generator  330 . Equation (18) depicts the function of generator  330 . Equations (38), (39) and (40) depict the function of processor  340 . The joint channel correlation generator  330  generates joint channel correlation matrix  334   1 - 334   4  between two transmit antennas and two receive antennas for even and odd sample streams. The joint channel correlation matrixes  334   1 - 334   4  are combined by a combiner  336  and the combined joint channel correlation matrix  338 , which corresponds to R fft  in Equation (42), enters a processor  340 .  
         [0074]     The processor  340  receives as an input the combined joint channel correlation matrix  338  and two combined results y (1) , y (2) ,  324   1 ,  324   2  and generates estimates of the transmitted chip sequences by solving the 2×2 linear systems of Equation (45). The equalized chip sequences  342   1 ,  342   2  are STTD decoded and soft combined by a STTD decoder/soft combiner  350  as shown in Equation (34). The STTD decoded and combined chip sequences  352  is processed by an IFFT unit  354  and despreader  356  to generate an estimate of transmitted data  358 .  
         [0075]     Although the features and elements of the present invention are described in the preferred embodiments in particular combinations, each feature or element can be used alone without the other features and elements of the preferred embodiments or in various combinations with or without other features and elements of the present invention.