Abstract:
A system and method for detecting, measuring, and reporting a time derivate of a current signal (di/dt) is provided. A sensing element detects current from a load. The sensing element includes an inductor. The inductor is located in series with the load and includes associated parasitic resistance. A differential potential develops across the inductor and the parasitic resistance. The differential potential is amplified and converted to a single-ended value. The single-ended value is then fed to an analog to digital converter that provides an output representative of di/dt.

Description:
RELATED APPLICATION  
       [0001]    This application claims priority to and incorporates by reference U.S. Provisional Application Ser. No. 60/311,653 filed Aug. 10, 2001. 
     
    
     
       FIELD  
         [0002]    The present invention relates generally to current sensors, and more particularly, relates to a current derivative sensor.  
         BACKGROUND  
         [0003]    The ability to detect, measure, and record a rate of current change may be critical in high-speed electronic applications. The rate of current change may be referred to as a slope of a current signal, or alternatively, as a time derivative of a current signal (di/dt). The rate of the current change may be important when accounting for unwanted noise, such as electromagnetic interference (EMI) and radio-frequency interference (RFI), generated by high-speed circuits. However, most current sensors detect and measure only the magnitude of the current, and not the rate of current change.  
           [0004]    Often the slope of the current signal is determined using computer simulation techniques, such as finite element analysis, lumped element simulation, and behavioral modeling. These computer techniques can become computationally intensive and include limiting assumptions, which may reduce the accuracy of the simulation result. Therefore, it would be beneficial to make a direct measurement of di/dt using a current derivative sensor.  
         SUMMARY  
         [0005]    A current derivative sensor and a method for detecting, measuring, and recording a time derivative of a current signal (di/dt) are provided. A sensing element detects current. The current flows through the sensing element, generating a differential potential across the sensing element. A gain circuit amplifies and converts the differential potential to a single-ended output. An analog to digital converter converts the single-ended output and provides an output representative of di/dt.  
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0006]    Presently preferred embodiments are described below in conjunction with the appended drawing figures, wherein like reference numerals refer to like elements in the various figures, and wherein:  
         [0007]    [0007]FIG. 1 is a circuit diagram of a current derivative sensor, according to an exemplary embodiment;  
         [0008]    [0008]FIG. 2A is a circuit diagram of a current derivative sensor, according to another exemplary embodiment;  
         [0009]    [0009]FIG. 2B is a circuit diagram of a current derivative sensor, according to another exemplary embodiment;  
         [0010]    [0010]FIG. 3A is a circuit diagram of a calibration circuit, according to an exemplary embodiment;  
         [0011]    [0011]FIG. 3B is a circuit diagram of a calibration circuit, according to another exemplary embodiment;  
         [0012]    [0012]FIG. 4 is a graph of selectivity of the sensor, according to an exemplary embodiment; and  
         [0013]    [0013]FIG. 5 is a flow chart diagram of a method of measuring a time derivative of a current signal (di/dt), according to an exemplary embodiment.  
     
    
     DETAILED DESCRIPTION  
       [0014]    [0014]FIG. 1 is a circuit diagram of a current derivative sensor  100 , according to an exemplary embodiment. The current derivative sensor  100  may include a sensing element  110 , a gain circuit  118 , an analog to digital (AID) converter  120 , and a calibration circuit  122 . The sensor  100  may be a standalone device that can be inserted into a desired circuit to make measurements, or it may be fabricated on a die and interconnected to circuits on a separate die, or may be fabricated on the same die as the circuit being measured.  
         [0015]    The current derivative sensor  100  may be designed to monitor current  102 . The current  102  may be composed of both a DC component and a transient component. The current  102  may be generated by a load  104 . The load  104  may be any conductor that can generate a transient current signal, such as integrated circuit interconnect metallization, integrated circuit polysilicon, silicided silicon connectors, printed circuit board traces, insulated wires, and non-insulated wires. For the sake of simplicity, the load  104  is depicted in FIG. 1 as a parallel combination of a capacitor  106  and a variable resistor  108 .  
         [0016]    The current  102  may be detected by the sensing element  110 . The sensing element  110  preferably includes an inductor  112  placed in series with the load  104 . A parasitic resistance may be associated with the inductor  112 . The parasitic resistance is depicted in FIG. 1 as a resistor  114  located in series with the inductor  112 . Other devices may also be included within the sensing element  110 .  
         [0017]    The current  102  may cause a differential potential  116  to develop across the series combination of the inductor  112  and the resistor  114  between the electrical nodes labeled in FIG. 1 as V positive  and V negative . The differential potential  116  may be generated from substantially three origins. A first origin may be a DC potential produced when the current  102  flows through the resistor  114 . A second origin may be a transient potential produced when the current  102  flows through the resistor  114 . A third origin may be transient potential produced when the current  102  flows through the inductor  112 . The differential potential  116  may be defined by the following equation:  
           v ( t )=( I+i ( t )) R+L ( di/dt )   Equation 1  
         [0018]    where: v(t) is the differential potential  116 ;  
         [0019]    I is the DC current flowing through the resistor  114 ;  
         [0020]    i(t) is the transient current flowing through the resistor  114 ;  
         [0021]    R is the resistance from resistor  114 ;  
         [0022]    L is the inductance from inductor  112 ; and  
         [0023]    di/dt is the transient current flowing through the inductor  112 .  
         [0024]    The current derivative sensor  100  may be designed to measure substantially the third origin of the differential potential  116 , which is the transient current di/dt flowing through the inductor  112 .  
         [0025]    The calibration circuit  122  may be used to calibrate the current derivative sensor  100 . The calibration circuit  122  may be used to measure the parasitic resistance. Additionally, the calibration circuit  122  may be used to determine an accurate value of inductance of the inductor  112 , which may be needed to correlate the differential potential  116  to the magnitude of the di/dt event. The details of the calibration circuit  122  are discussed below.  
         [0026]    The gain circuit  118  may be used to amplify and convert the differential potential  116  to a single-ended output. The gain circuit  118  may be a differential-input, single-ended output operational amplifier (op amp). The addition of single-ended gain circuits  202  located at an output of the op amp may be beneficial for amplifying small differential potentials, as shown in FIG. 2A. Alternatively, additional differential-input, single-ended output gain circuits  204  located at the output of the op amp may be used to provide additional amplification for small differential potentials, as shown in FIG. 2B.  
         [0027]    Referring back to FIG. 1, an output of the gain circuit  118  may be connected to the A/D converter  120 , one embodiment of which is shown as a plurality of switches. While Schmitt triggers are used in a preferred embodiment, other switching devices or combination of devices that can be triggered may also be employed.  
         [0028]    The Schmitt triggers may be configured such that LOW-to-HIGH input transition voltages are monotonically increasing from substantially a ground potential to a maximum supply voltage. The LOW-to-HIGH input transition voltage may be a value of voltage that causes a switch to change states from off to on. Alternatively, the LOW-to-HIGH input transition voltages may be monotonically increasing for a range of voltages located between the ground potential and the maximum supply voltage.  
         [0029]    Additionally, the Schmitt triggers may be configured such that HIGH-to-LOW input transition voltages are substantially at the maximum supply voltage. The HIGH-to-LOW input transition voltage may be a value of voltage that causes a switch to change states from on to off.  
         [0030]    In further alternative embodiments, the A/D converter  120  may be implemented using a series of voltage comparators having different reference voltages. Other suitable alternative analog to digital conversion techniques and circuits may also be used.  
         [0031]    When a small di/dt event occurs, only the Schmitt triggers with a LOW-to-HIGH input transition voltage near the ground potential may change to a HIGH output. A larger di/dt event may cause an increasing number of the Schmitt triggers to change to a HIGH output. A change in state to a HIGH output may be maintained until the A/D converter  120  is reset.  
         [0032]    Before being reset, the output of the A/D converter  120  may be detected providing an actual measurement of di/dt. The output of the A/D converter  120  may be displayed on a thermometer-type scale readout. (The thermometer-type scale readout is not shown in FIG. 1.) For example, if none of the switches have changed to a HIGH output, then the readout may be substantially zero or at the bottom of the scale. As the number of switches that have changed to a HIGH output increases, the readout may be increased accordingly up the scale. When all of the switches have changed to a HIGH output, the readout may be substantially at full scale. This scale readout may be recorded. While the thermometer-type scale readout is used in a preferred embodiment, other methods of displaying the output of the A/D converter  120  may also be employed.  
         [0033]    In an alternative embodiment, the switch values may be converted to a binary number representing the number of switches in the HIGH state. For example, if there are seven switches, and a given output of gain circuit  118  causes the first five switches to change state (e.g., to a HIGH output from a LOW output), then the switch outputs may be converter by an encoder circuit (not shown) into the binary value “101”. A suitable encoder circuit may be implemented using a digital counter to count the number of switch outputs in the HIGH state, a look-up table, a combinational logic circuit, etc.  
         [0034]    Alternatively, the output of the A/D converter  120  may correspond to a memory address, such as a Read-Only-Memory (ROM) address. For this example, “101” may be stored in a ROM address corresponding to the first five switches being in the HIGH state.  
         [0035]    [0035]FIG. 3A is a circuit diagram of a calibration circuit  300 , according to an exemplary embodiment. The calibration circuit  300  is substantially the same as the calibration circuit  122  as shown in FIG. 1. The calibration circuit  300  includes a plurality of precision matched current sources  302  of substantially the same DC magnitude, I. The plurality of current sources  302  may be connected in parallel through independently controlled switches  304 . FIG. 3A depicts the switches as field-effect transistors; however, other switches may be used. The independently controlled switches  304  may be controlled by a variety of devices, such as a microcontroller.  
         [0036]    Activating a single leg of the parallel network of current sources  302  by closing one or more of the switches  304  may generate a known value of DC current when the current through the inductor  112  has reached steady state. Using the known value of DC current, a differential potential  116  may develop across resistor  114  and be amplified by the gain circuit  118 . The AID converter  120  may detect the amplified differential potential signal. The output of the A/D converter  120  may be used to determine the value of the parasitic resistance, depicted in FIG. 1 as resistor  114 .  
         [0037]    Using the value of the parasitic resistance, the calibration circuit  300  may determine an accurate value of inductance of inductor  112 , which may be needed to correlate the differential potential  116  to the magnitude of the di/dt event. A current with a known di/dt may be generated by incrementally activating successive legs of the parallel network of current sources  302  by closing the switches  304  one at a time. With each successive activation, the current generated may be increased by substantially the DC magnitude, I, of the current sources  302 . Because the calibration circuit  300  is able to account for the contribution of the parasitic resistance, the circuit may accurately determine, or measure, the inductance. As seen with reference to Equation 1, the inductance is equal to the difference between the differential potential  116  and the contribution of the parasitic resistance, divided by the known di/dt.  
         [0038]    A clock circuit or a timer may be used to control the successive activation of the legs of the parallel network of current sources  302 . (The clock circuit and timer are not shown in FIG. 3A.) As shown in FIG. 3B, a filter  306  may be added to the calibration circuit  300 , as the current waveform generated will have a staircase response. For example, the filter  306  may be a low pass filter. The addition of the low pass filter  306  may smooth the waveform, which may be a closer approximation of a ramp with a constant di/dt. Alternatively, because the ramp response ideally contains only odd-order harmonics, a filter  306  operable to remove even-order harmonics may be added to the calibration circuit  300 .  
         [0039]    In an alternative embodiment, the calibration circuit  300  may include a control circuit. (The control circuit is not shown in FIG. 1.) The control circuit may include a microcontroller. Alternatively, the control circuit may include a logic circuit providing combinational and/or sequential logic. For example, the logic circuit may be a state machine.  
         [0040]    The control circuit may be operable to provide the known value of DC current by controlling the operation of independently controlled switches  304 . The control circuit may open and close switches  304  based on what type of calibration is being performed.  
         [0041]    In addition, the control circuit may receive voltage information from the output of the gain circuit  118  and/or the A/D converter  120 . The control circuit may receive the voltage information for various DC current values. For example, the control circuit may receive the voltage information for different combinations of switches  304  being opened and closed. The control circuit may store the voltage information for the various DC current values.  
         [0042]    When the current derivative sensor  100  is operating, the control circuit may receive the voltage information from the output of the gain circuit  118  and/or the A/D converter  120  and subtract the previously stored voltage information for the corresponding DC level. As such, the control circuit may provide an output signal that has been compensated for the DC component of current.  
         [0043]    Additionally, the control circuit may provide an offset value based on the DC component of current to the gain circuit  118 . Furthermore, for the A/D embodiment using voltage comparators, the control circuit may provide the different reference voltages to the voltage comparators. The different reference voltages may or may not be linearly spaced.  
         [0044]    To maximize the sensitivity of the current derivative sensor  100 , the inductive component of the differential potential  116  may be emphasized, minimizing the resistive component. Referring back to Equation  1 , to emphasize the inductive component, the following equation holds true.  
           L ( di/dt )&gt;&gt;( I+i ( t )) R    Equation 2  
         [0045]    The inductive component is related to the quality factor Q of the inductor. The quality factor can be defined as:  
           Q=ωL/R= 2π fL/R    Equation 3  
         [0046]    where: ω is the angular frequency of an AC signal and f is the equivalent frequency in Hertz. If the AC signal is represented by only its fundamental frequency then:  
           di/dt= 2 f.    Equation 4  
         [0047]    Combining Equations 2, 3, and 4 yields the following design equation:  
           Q&gt;&gt;π (1+ I/i ).   Equation 5  
         [0048]    The significance of the design equation, Equation 5, on the performance of the sensor is shown graphically in FIG. 4. FIG. 4 depicts the selectivity of the current derivative sensor  100 . The selectivity of the current derivative sensor  100  may be defined as the percentage of the differential potential  116  due to the inductive contribution, L(di/dt).  
         [0049]    A ratio is defined between the DC current I and the transient current i. A high ratio may imply that the total current  102  is nearly constant with small transient variations. In this situation, the inductive contribution would not be emphasized, and the selectivity would be low. This may cause the scaled output to be largely a result of the parasitic resistance. On the other hand, a low ratio may imply that the current  102  is dominated by the transient current i.  
         [0050]    The impact of the quality factor is also depicted in FIG. 4. A larger value of Q results in a higher selectivity of the inductive contribution. A high value of Q may be indicative of either a high value of di/dt or a low value of resistance  114 . Conversely, a low value of Q may be indicative of either a low value of di/dt or a high value of resistance  114 .  
         [0051]    [0051]FIG. 5 depicts a flow chart diagram of a method  500  of measuring a time derivative of a current signal (di/dt). Step  502  is sensing the current. The sensing element  110  may be used to sense the current  102  generated by the load  104 . The differential potential  116  may be generated as the current  102  flows through the sensing element  110 .  
         [0052]    Step  504  is amplifying the differential potential  116 . The gain circuit  118  may be used to amplify the differential potential  116 . The gain circuit  118  may include additional gain circuits  202 ,  204  to provide more amplification for small differential potentials. In an alternative embodiment, results from the calibration circuit  300 , such as the inductance value, may be used to set the gain of the gain circuit  118 .  
         [0053]    Step  506  is converting the differential potential  116  to a singled-ended output. This step may be accomplished using the same gain circuit  118  used for amplification in Step  504 .  
         [0054]    Step  508  is triggering the A/D converter. The A/D converter may be a plurality of Schmitt triggers configured such that the input transition voltage operable to trigger a switch from off to on is monotonically increasing from switch to switch. Only the switches with transition voltages at or below the voltage applied by the gain circuit  118  will turn on. The number of switches that turn on for a given period of time is representative of the time derivative of the current signal  102 . The greater the value of voltage at the output of the gain circuit  118 , the greater the number of switches that will turn on, which represents a greater rate of current change.  
         [0055]    The output of the A/D converter  120  may be displayed on a thermometer-type scale readout, which may then be recorded. In an alternative embodiment, the switch values may be converted to a binary number representing the number of switches in the HIGH state as previously discussed.  
         [0056]    The current derivative sensor  100  may be able to detect electromagnetic interference and/or radio-frequency interference. For example, the current derivative sensor  100  may be able to detect di/dt events ranging from 10 3  amps/second (or 1 amp/millisecond) to 10 12  amps/second (or 1 amp/picosecond). However, other results may be possible based on the effects of the parasitic resistance depicted in FIG. 1 as resistor  114 , the quality factor of the inductor  112 , and a limit on the gain circuit  118 , which may be imposed by the maximum supply voltage. The ability to measure such a wide range of di/dt events makes the current derivative sensor  100  ideally suited for applications involving integrated circuits, printed circuit boards, insulated and non-insulated wiring, and other electrical conductors.  
         [0057]    It should be understood that the illustrated embodiments are exemplary only and should not be taken as limiting the scope of the present invention. The claims should not be read as limited to the described order or elements unless stated to that effect. Therefore, all embodiments that come within the scope and spirit of the following claims and equivalents thereto are claimed as the invention.