Abstract:
An efficient, low-loss, low sidelobe, high dynamic range phased-array radar antenna system is disclosed that uses metamaterials, which are manmade composite materials having a negative index of refraction, to create a biconcave lens architecture (instead of the aforementioned biconvex lens) for focusing the microwaves transmitted by the antenna. Accordingly, the sidelobes of the antenna are reduced. Attenuation across microstrip transmission lines may be reduced by using low loss transmission lines that are suspended above a ground plane a predetermined distance in a way such they are not in contact with a solid substrate. By suspending the microstrip transmission lines in this manner, dielectric signal loss is reduced significantly, thus resulting in a less-attenuated signal at its destination.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application claims priority to U.S. Provisional Patent Application, Ser. No. 60/550,473, entitled Phased Array Metamaterial Antenna System, filed Mar. 5, 2004. 
    
    
     FIELD OF THE INVENTION 
     The present invention relates to phased array antenna systems and, more particularly, to phased array antenna systems useful in automotive radar applications. 
     BACKGROUND OF THE INVENTION 
     Phased array systems and antennas for use in such systems are well known in, for example, telecommunications and radar applications. Such systems generally employ fixed, planar arrays of individual transmit and receive elements. When receiving electromagnetic (EM) signals, such as a communication signal or the return signal in a radar system, phased array systems receive signals at the individual elements and coherently reassemble the signals over the entire array by compensating for the relative phases and time delays between the elements. When transmitting signals, beams are electronically steered by delaying the excitation of selected individual radiating elements. For relatively small antennas, adequate delays of the individual elements can be provided by adjusting the phase of the excitation signals supplied to the elements. 
     Traditional phased-array antenna systems used in such applications were expensive to manufacture, were relatively large and bulky, and the performance was less than desirable due to, for example, relatively poor performance of monolithic microwave integrated circuits (MMICs) of the transceiver section of the antenna system. For example, such MMICs typically resulted in significant undesirable sidelobes which limited the usefulness of antennas using such circuits. Recent attempts at such antenna systems have included printing antenna system elements, such as signal traces and patch antennas, on a circuit board using well-known lithography techniques. Such antenna systems solve one problem in that they are smaller and relatively inexpensive to manufacture and, therefore, have been used increasingly in new applications. One such application is in adaptive cruise control systems in trucks, automobiles and other such vehicles. Such cruise control systems are able to reduce or increase the speed of the vehicle in order to maintain a predetermined distance between the vehicle and other traffic. Radar systems in vehicles are potentially also useful in such applications as collision avoidance and warning. 
     SUMMARY OF THE INVENTION 
     The present inventor has realized that, while the size and cost of in-vehicle phased array antenna systems has improved, due in part to the lithographic processes used to manufacture modern antenna systems, even the improved antenna systems are limited in certain regards. For example, recent attempts of implementing in-vehicle radar have focused on the 76–77 GHz frequency range and recent data communications attempts have been made in the 71–76 GHz and the 81–86 GHz frequency range. However, at such frequencies, antenna systems with lithographically-printed microstrip transmission lines experience a high degree of signal attenuation. Additionally, such printed antenna systems have relied on a signal-feed/delay line architecture that resulted in a biconvex, or Fresnel, lens for focusing the microwaves. The use of such lens architectures resulted in microwave radiation patterns having poor sidelobe performance due to signal attenuation of electromagnetic energy as it passed through the lens. Specifically, the signal passing through the center portion of the lens was attenuated to a greater degree than the signal passing through the edges of the lens, thus resulting in significant sidelobes. While signal delay lines in the lens portion of the system could reduce the sidelobes and, as a result, increase the amplitude performance of the phased array system, this was also limited in its usefulness because, by implementing such delay lines, the operating bandwidth of the phased-array system was reduced. 
     Therefore, the present inventor has invented an efficient, low-loss, low sidelobe, high dynamic range phased-array radar antenna system that essentially solves the aforementioned problems. In one embodiment, the present invention uses metamaterials, which are manmade composite materials having a negative index of refraction, to create a biconcave lens architecture (instead of the aforementioned biconvex lens) for focusing the microwaves transmitted by the antenna. Accordingly, a signal passing through the center of the lens is attenuated to a lesser degree relative to the edges of the lens, thus significantly reducing the amplitude of the sidelobes of the antenna while, at the same time, retaining a relatively wide useful bandwidth. 
     In another embodiment, attenuation across microstrip transmission lines is reduced by using low loss transmission lines that are suspended above a ground plane a predetermined distance in a way such they are not in contact with a solid substrate. By suspending the microstrip transmission lines in this manner, dielectric signal loss is reduced significantly, thus resulting in a less-attenuated signal at its destination. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWING 
         FIG. 1  shows a prior art monolithic microwave integrated circuit phased-array antenna system; 
         FIG. 2  shows how the antenna system of  FIG. 1  can be used to transmit an electromagnetic signal; 
         FIGS. 3A and 3B  show how an electromagnetic signal radiated by the system of  FIG. 1  can be steered in different directions by selecting an appropriate signal input line; 
         FIGS. 4A and 4B  show illustrative metamaterials useful in the electromagnetic lens portion of the system of  FIG. 1 ; and 
         FIG. 5  shows a suspended transmission line. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       FIG. 1  shows one illustrative, relatively low-cost prior art antenna system potentially useful for telecommunications and in-vehicle radar uses. Specifically,  FIG. 1  shows a monolithic microwave integrated circuit (MMIC) phased array antenna system  100  which has antenna  101 , lens portion  102 , waveguide  103  and signal input lines  150 – 158 . Antenna  101  has an array of antenna elements  101  wherein the individual elements  104  of each column  105  are electrically connected to each other. The individual columns  105  are, for example, lithographically printed microstrip lines with printed antenna patches disposed periodically along the microstrip lines. Each column  105  of antenna elements  104  is connected to one of delay lines  107  which are suitable for use as waveguides for electromagnetic signals. Delay lines  107  are, for example, microstrip lines lithographically printed on a suitable substrate. One or more electronic components, such as amplifiers, may be disposed along each of the delay lines  107 . Delay lines  107  form lens  102  which is an electromagnetic lens that is used to delay and/or amplify the individual signals traveling across each delay line. Such delay lines are used in order to compensate for the aforementioned poor sidelobe performance of traditional Fresnel or biconvex lenses. As is well known, such delays serve to excite the individual antenna elements  104  at desired times relative to the other antenna elements in antenna  101  to steer and focus the radio frequency beams produced by antenna  101 . However, as one skilled in the art will recognize, delay lines  107  also reduce the useful bandwidth of the phased array antenna system. 
     Waveguide  103  is, illustratively, a parallel plate wave guide printed lithographically on a suitable dielectric substrate. Such lithographic processes are well known in the art. Waveguide  103  functions to receive signals from any of signal input lines  150 – 158  and to guide those signals in a predetermined fashion to the individual delay lines  107  of lens  102 . Signal input lines  150 – 158  are, for example, lines connected to a radar signal generating and processing system. 
       FIG. 2  shows how waveguide  103  functions to guide signals to delay lines  107 . Specifically, when the radar generating and processing system connected to signal input lines  150 – 158  generates a radar signal  203  for transmission, it transmits the signal across one or more of the input lines  150 – 158 , here, illustratively, input signal line  154 . When signal  202  reaches waveguide  103 , wavefront  201  spreads and propagates across the wave guide in direction  204  toward delay lines  107 /lens  102 . Thus, when wavefront  201  reaches the delay lines  107 , the signal will enter each delay line at substantially the same time with substantially the same phase. In the embodiment of  FIG. 2 , when a signal is transmitted across signal line  154 , the transmitted beam  203  is perpendicular to the face of antenna  101 . The lengths of delay lines  107  are chosen in a way such that sidelobes are reduced (relative to a Fresnel or biconvex lens without such lines) and a desirable beam amplitude profile is achieved. 
     It will be apparent to one skilled in the art that, in order to steer and focus the beam in the correct direction, the radar signal generating and processing system can transmit the signal across a different one or more of the signal input lines  150 – 158 . For example, referring to  FIG. 3A , if signal  302  is introduced to signal input line  158 , when it reaches waveguide  103  wavefront  301  will be created traveling in direction  303  across the waveguide. The signal will first reach the delay line  309  corresponding to column  310  of individual elements. The signal will progressively travel across the waveguide sequentially reaching delay lines in the plurality of delay lines  102  with a slightly delayed phase relative to the signal traveling across delay line  309 . As a result, it will be clear to one skilled in the art that the signal transmitted by antenna  101  will be steered in, for example, direction  304 . Likewise, referring to  FIG. 3B , by introducing a signal into signal input line  150 , wave front  305  will travel across the waveguide  103  in direction  307 , first reaching delay line  311  corresponding to column  312  of antenna elements. Accordingly, the signal transmitted by the antenna is steered in, for example, direction  308 . 
     While the MMIC prior art antenna structures of  FIGS. 1 ,  2 ,  3 A and  3 B are useful in many regards, they are limited in certain respects. For example, as discussed previously, delay lines  107  function to achieve a desirable signal amplitude profile with low sidelobes for a beam transmitted by antenna  101 . However, MMIC antennas using a lens structure such as lens structure  102  in  FIG. 1  can be relatively poor performing in terms of useable bandwidth and undesirably high sidelobes may still result. 
     Instead of using a biconvex lens structure, therefore, the present inventor has recognized that it would be desirable to use a biconcave lens structure that would result in lower attenuation at the center of the lens than at the edges and, as a result, result in a desirable amplitude profile of the transmitted beam without using bandwidth-limiting delay lines. However, to date, such a concave lens architecture has been difficult to achieve with conventional materials because naturally-occurring materials typically have a positive index of refraction and, hence, a biconcave lens made of such material would scatter, and not focus, light. However, recent material advances in composite structures known as metamaterials has introduced new physical structures with unique properties. The present inventor has realized that, by integrating metamaterials into the delay lines  107  of the lens portion  102  of  FIG. 1 , a biconcave lens structure can be achieved. 
     A great deal of recent research has been accomplished on the manufacture, properties and uses of metamaterials. Metamaterials, as used herein, are man-made composite structures that are characterized by a negative permittivity and a negative permeability at least across a portion of the electromagnetic frequency spectrum. Accordingly, the refractive index of a metamaterial is also negative across that portion of the spectrum. In practical terms, materials possessing such a negative index of refraction are capable of refracting propagating electromagnetic waves incident upon the metamaterial in an opposite direction compared to if the wave was incident upon a material having a positive index of refraction. If the wavelength of the electromagnetic energy is relatively large compared to the individual structure elements of the metamaterial, then the electromagnetic energy will respond as if the metamaterial is actually a homogeneous material. 
       FIGS. 4A and 4B  show a top view and a three dimensional view of illustrative metamaterial structures that are useful in accordance with the principles of the present invention in the antenna structure of  FIG. 1 . The metamaterials of  FIGS. 4A and 4B  are illustratively of the type investigated by Christophe Caloz et al. of the University of California, Los Angeles, Department of Electrical Engineering. Examples of the principles underlying such metamaterials can be found in  Microwave Circuits Based on Negative Regractince Index Material Structures , Caloz et al., 33 rd  European Microwave Conference, conference report, p. 105, Munich, Germany 2003;  Positive/Negative Refractice Index Anisotropic  2- D Metamaterials , Caloz et al, IEEE Microwave and Wireless Components Letters, Vol. 13, No. 12, p. 547, December 2003 ; Invited—Novel Microwave Devices and Structures Based on the Transmission Line Approach of Meta - Materials , Caloz et al., 2003 IEEE MTT-S Digest, p. 195 ; A Broadband Left - Handed  ( LH )  Coupled-Line Backward Coupler with Arbitrary Coupling Level , Caloz et al., 2003 IEEE MTT-S Digest, p. 317; and  A Novel Mixed Conventional MIcrostrip and Composite Right Left - Handed Backward - Wave Directional Coupler With Broadband and Tight Coupling Characteristics , Caloz et. al., IEEE Microwave and Wireless Components Letters, Vol. 14, No. 1, January 2004, p. 31. Each of the foregoing publications are hereby incorporated by reference herein in their entirety. 
     Referring to  FIG. 4A , structure  400  is an illustrative microstrip line  401  developed by Caloz et al., wherein a plurality of unit-cell circuit structures are repeated periodically along the microstrip line. A unit-cell circuit structure merely is one or more electrical components, in this case disposed along the microstrip transmission line. In  FIG. 4A , for example, series interdigital capacitors  402  are placed periodically along the line  401  and T-junctions  403  between each of the capacitors  402  connect the microstrip line  401  to shorted spiral stub delay lines  404  that are, in turn, connected to ground by vias  405 . The microstrip structure of one of the aforementioned capacitors, one spiral inductor, and the associated ground via, forms the unit-cell circuit structure of  FIG. 4A . By using a plurality of microstrip lines in place of the delay lines  107  in  FIG. 1 , the phases of the signals traveling along the edges of the lens are delayed relative to those traveling in the center of the lens. Thus, the amplitude of the center portion of the beam transmitted by antenna  101  is higher than the amplitude at the edges and, accordingly, sidelobes are reduced. One skilled in the art will recognize that other suitable unit-cell circuit architectures may be used to achieve the propagation characteristics useful in accordance with the principles of the present invention. For example,  FIG. 4B  shows a 3 dimensional representation of a microstrip metamaterial structure that does not rely on spiral inductors. 
     Caloz reported in the publication  Invited—Novel Microwave Devices and Structures Based on the Transmission Line Approach of Meta - Materials  referenced above, that structures similar to  FIG. 4A  could be used in leaky wave antennas (not phased array antennas) that were designed to operate at frequencies up to approximately 6.0 GHz. The present inventors, however, have realized that, with certain modifications, these metamaterials can be used at relatively high frequencies, such as those frequencies useful in automotive radar and/or data communications applications above 60 GHz and, more particularly, between 76 GHz and 77 GHz (for automotive radar) and 71–76 and 81–86 GHz (for data communications). For example, the unit cell-circuit structure of  FIG. 4A  can be reduced to a size smaller than the wavelength of the signal. It is obvious to one skilled in the art, in light of the teachings herein, how to design the metamaterial microstrip line (e.g., physical size and positioning of unit cells) to achieve a desired transmission line impedance at a particular frequency. 
     One problem with using the above-described metamaterial structures in high-frequency applications is that such high-frequency signals traveling across microstrip lines experience a high degree of attenuation. Specifically, as frequencies rise to ≧70 GHz, signal attenuation for a given traditionally-designed transmission line length increases significantly and, accordingly, the received signal strength at a signal&#39;s destination is significantly reduced. Thus, traditional microstrip transmission lines are inadequate for use at such high frequencies. Such signal attenuation and methods for reducing the attenuation is the subject of copending U.S. patent application Ser. No. 10/788,826, entitled Low-Loss Transmission Line Structure, filed Feb. 27, 2004. This patent application is hereby incorporated by reference herein in its entirety. 
     As discussed more fully in the 10/788,826 application,  FIG. 5  shows one illustrative embodiment of a transmission line structure  500  in accordance with the principles of the present invention whereby the aforementioned dielectric signal loss is reduced or substantially eliminated. Specifically,  FIG. 5  shows an illustrative transmission line  501  that is physically suspended above substrate  502  which is, illustratively, a metallized layer functioning as an electrical ground for transmission line  501 . Transmission line  501  is also referred to herein interchangeably as a transmission element. One skilled in the art will recognize that substrate  502  may be, for example, a layer of gold, copper, aluminum, or another electrically conducting material suitable for use as a ground plane. Support elements  503 , here illustratively bent support arms, are attached to both the transmission line and the substrate and function to both support the transmission line above the ground substrate  502  as well as, illustratively, to electrically connect the transmission line to that substrate. Once again, support arms  503  may be, illustratively, manufactured from an electrically conducting material such as the aforementioned gold, copper or aluminum or any other electrically conducting material. One skilled in the art will recognize that other materials, such as plastic may be used to support the transmission element. Support arms  503  have length L and height H and are spaced a distance D from each other. One skilled in the art will recognize that L, D and H can be selected to produce a desired electrical property of transmission element  501 , such as the impedance of the transmission line. For example, if the line width W is selected as 1.08 mm, the length L of the support arms is selected as 3.01 mm, the height H is selected as 250 micrometers, and the support arms are separated by 4 mm from each other, transmission line  501  will illustratively have approximately a 50 Ohm impedance, which is desirable in a number of applications. Other dimensions may be selected to produce a variety of desirable transmission line impedances. The transmission line structure  500  of  FIG. 5  substantially reduces the signal attenuation of a high-frequency RF signal propagating along transmission line  201 . This reduction is the result of separating the transmission line from the substrate and, accordingly, reducing the exposure of the propagating signal to any electromagnetic field present in the substrate. One skilled in the art will fully recognize that, by applying the above-described method to suspend a transmission line above the associated ground plane, attenuation in the metamaterial structures of  FIGS. 4A and 4B  can be significantly reduced or eliminated. 
     The foregoing merely illustrates the principles of the invention. It will thus be appreciated that those skilled in the art will be able to devise various arrangements which, although not explicitly described or shown herein, embody the principles of the invention and are within its spirit and scope. Furthermore, all examples and conditional language recited herein are intended expressly to be only for pedagogical purposes to aid the reader in understanding the principles of the invention and are to be construed as being without limitation to such specifically recited examples and conditions. Moreover, all statements herein reciting aspects and embodiments of the invention, as well as specific examples thereof, are intended to encompass functional equivalents thereof.