Abstract:
The invention relates to a novel ballast for a discharge lamp, in which a continuous-operation control circuit is brought out of operation when a physical operation, which defines a preheating time, of a preheating timer has as yet not been sufficiently returned if the lamp is intended to be restarted.

Description:
FIELD OF THE INVENTION 
   The present invention relates to a ballast for discharge lamps, to be precise specifically those discharge lamps which have preheatable electrodes. 
   BACKGROUND OF THE INVENTION 
   Such ballasts are known per se. They frequently have half-bridge inverter circuits. However, the invention also relates to other ballasts. In principle, an inverter circuit generates from a rectified AC voltage supply or a DC voltage supply a supply power for the lamp which has a higher frequency than the system frequency. In many cases, a control circuit is provided here for controlling the lamp current or the lamp power during continuous operation of the lamp, and this will be referred to below as the continuous-operation control circuit. This continuous-operation control circuit influences the operating frequency at which the inverter supplies power to the lamp and thereby controls the lamp current or the lamp power. This takes place by bringing the operating frequency closer to or further away from resonant frequencies of lamp resonant circuits containing the lamp. 
   Before the lamp can be operated, it has to be started by a relatively high voltage. For this purpose too, resonance excitation of the lamp resonant circuit is used in many cases. In the case of discharge lamps having preheatable electrodes, the electrodes are initially preheated for a specific time before the actual starting voltage is applied. The preheating time is in this case determined by a preheating timer, in which, in the most general sense, a physical operation runs which defines a temporal delay, and, once the preheating time has expired, must return in order to be able to run again for subsequently switching the lamp on again. The preheating timer in this case has the function of a switch. The details on the implementation of such a preheating timer and the physical operation are not relevant to the principle of the invention, for which reason the abovementioned general wording has been selected. 
   In such cases, starting of the lamp takes place independently of the continuous-operation control circuit once said physical operation has run. For this purpose, in some way the starting voltage must be reached, for example by resonance excitation in the lamp resonant circuit. In this case the influence of the continuous-operation control circuit would have a disruptive effect. 
   SUMMARY OF THE INVENTION 
   The invention is based on the technical problem of specifying an improved ballast and an improved operating method for discharge lamps having preheatable electrodes using a continuous-operation control circuit. 
   It relates to an electronic ballast for at least one discharge lamp having preheatable electrodes, which ballast has a continuous-operation control circuit for controlling the lamp current or the lamp power during continuous operation of the lamp via the operating frequency of the lamp, a preheating timer, which defines a preheating time for the electrodes and is designed to define the preheating time by means of a physical operation which runs with a temporal delay and then to allow this operation to return with a temporal delay, the ballast being designed to start the lamp independently of the continuous-operation control circuit when the physical operation of the preheating timer has run, characterized in that the ballast is also designed to bring the continuous-operation control circuit for continuous operation of the lamp out of operation when the preheating element, once operation of the lamp has been interrupted owing to an as yet incomplete return of its physical operation, cannot define a complete new preheating operation, with the result that the lamp can then be started independently of the continuous-operation control circuit. 
   The invention is also based on a corresponding operating method. 
   The inventor has established the starting basis of the invention as being the fact that it is possible for problems to result from the temporal delays of the preheating timer. In general, the physical operations defining the preheating time also return again with a specific temporal delay. 
   This applies, for example, to the case, which is also preferred here, of a PTC thermistor, which is heated during the preheating time by resistive heat losses, as the preheating timer, the PTC thermistor in this case increasing its electrical resistance value as a result of the increasing temperature. An important mechanism which is preferred here is in this case damping of the lamp resonant circuit, which damping decreases with the increasing PTC thermistor value, and starting as a result of resonance excitation therein. If the PTC thermistor is now heated, it then cools down again only slowly. Even continuous heating of the PTC thermistor during continuous operation of the lamp is also to be expected, since small currents flow through it continuously. The cooling process thus begins only once the lamp has been switched off. In the case of the PTC thermistors used for electronic ballasts, the cooling process typically takes several tens of seconds to several minutes and is thus markedly slower than the typical cooling time of the electrodes of approximately several 100 ms. If the discharge lamp is thus switched on again after a relatively short period of time, the PTC thermistor has not sufficiently cooled down again or, in more general terms, the physical operation of the preheating timer has not returned to a sufficient extent. In such cases, operational faults may occur by the continuous-operation control circuit coming into operation or remaining in operation owing to the apparent expiry of the preheating time. This generally disrupts or prevents restarting of the lamp. 
   The above description would also apply in the same sense for the case in which the physical operation of the preheating timer returns as soon as during the continuous operation of the lamp, i.e. has returned after a relatively long operation. In this case, situations are nevertheless possible in which the lamp is switched on only briefly, is immediately switched off again and thereupon is switched on again relatively rapidly. For example, this may take place when a lamp, luminaire or illumination system is newly installed and where it is necessary for its operability to be “repeatedly tested”. In such cases, the operating personnel generally do not know the background of the failure to restart and consider the lamp or luminaire to be defective. 
   The invention therefore proposes bringing the continuous-operation control circuit out of operation for the case of a physical operation in the preheating timer which has not returned to a sufficient extent, in order to make it possible to restart independently of the continuous-operation control circuit. 
   This preferably takes place by the lamp voltage, a potential derived therefrom or another variable correlating therewith being applied to an input of a control amplifier or switching transistor in the continuous-operation control circuit. It may of course also be sufficient to merely use a time component of the continuous-operation control circuit or the correlating variable. Reference is made to the exemplary embodiments. 
   It has already been established above that a PTC thermistor is a common and in this case preferred preheating timer. However, in principle other preheating timers also come into consideration, in particular switches which can be driven by means of timers, for example RC elements. 
   For the case of a PTC thermistor, the invention also provides for a threshold value component to preferably be connected in series with the PTC thermistor, for example a so-called TISP or SIDAC, i.e. a threshold value component which does not conduct a current below a specific voltage threshold value. This provides the possibility, which has already been discussed at the outset, of the PTC thermistor, which is generally connected in parallel with the lamp, not conducting a current during continuous operation but only in the preheating and starting phases, during which higher voltages are applied. 
   It is generally necessary for a lamp current measurement to be provided for the continuous-operation control circuit either because the lamp current itself is controlled or because the lamp power is determined from the lamp current. In this case, the invention proposes different preferred variants. Firstly, the lamp current may be measured in series with a coupling capacitor which connects one of the lamp electrodes to one of the supply branches of the ballast. The term “coupling capacitor” generally refers to capacitors which are connected in series with the lamp or the lamps and which prevent a steady-state direct current through the lamp(s). 
   In this case, preferably with at least one pair of diodes, a branch is provided in which a measurement is carried out only during one half-cycle, and thus no energy is consumed during the other half-cycle. For this purpose, a current measuring resistor is connected in series with one of the diodes. Reference is made to the exemplary embodiments. 
   A likewise favorable solution which is, however, slightly more complex involves a measuring transformer. Preference is given in this case in particular to a differential current transformer, with which a correction can be made to the total lamp current by the preheating current or the current flowing through the electrodes and, for example, the PTC thermistor even during continuous operation. Only the current actually flowing through the discharge in the lamp is thus considered to be the lamp current. 
   A further, preferred refinement of the invention provides a voltage control circuit, which serves the purpose of adjusting the starting voltage of the lamp resonant circuit using the frequency of the half-bridge or another converter in the ballast. This voltage control circuit is advantageous since, when starting using resonance excitation as a result of the required magnification factor of the lamp resonant circuit, a relatively accurate frequency adjustment is required. The control circuit can in this case match the frequency to the resonance response of the lamp resonant circuit and can in this case operate in particular by means of limiting the starting voltage by altering the frequency. 
   The abovementioned continuous-operation control circuit may be combined with the voltage control circuit to such an extent that both have access to the same control input for controlling the operating frequency of the converter. In this case, provision may preferably be made for the circuit to function as a current or power control circuit (i.e. continuous-operation control circuit) as soon as notable lamp currents flow, i.e. the lamp has been started, and, in the other case, the voltage control “has priority”. The abovementioned consideration of the preheating current or PTC thermistor current in the lamp current measurement is of importance here. However, it is also possible for a realistic lamp current measurement to be undertaken without a differential current transformer, for example by the current control being blocked during the preheating phase by a voltage measurement via the PTC thermistor (or else via a measuring resistor in parallel or in series with the PTC thermistor). 
   In many cases, ballasts are designed to operate a plurality of lamps. If these lamps are connected in series, no significant additions need to be made to the abovementioned designs, as is shown in the corresponding exemplary embodiment. If they are connected in parallel, it is particularly expedient to connect the corresponding lamp voltages or variables correlating therewith to the input of the control amplifier or switching transistor in the continuous-operation control circuit in the form of an exclusive-OR combination. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The invention will be explained in more detail below with reference to three exemplary embodiments. The individual features disclosed therein may also be essential to the invention in other combinations. The description above and below relates to the apparatus aspect and the method aspect of the invention without this being explicitly mentioned in detail. 
       FIG. 1  shows a circuit diagram relating to a first exemplary embodiment according to the invention. 
       FIG. 2  shows a circuit diagram relating to a second exemplary embodiment according to the invention. 
       FIG. 3  shows a circuit diagram relating to a third exemplary embodiment according to the invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1  shows a first exemplary embodiment. Shown on the left are two connections KL 1 - 1  and KL 1 - 2 , to which a system voltage can be connected. A filter comprising two capacitors C 1  and C 2  and two coupled coils, designated FI 1 , connects the system voltage connections to a full-bridge rectifier comprising the diodes D 1 -D 4 . The rectified supply voltage is connected to an intermediate circuit storage capacitor C 6 , shown on the very right in the figure, via diodes D 5 -D 8  which are to be considered as two pump branches. 
   In order to adhere to relevant specifications as regards system current harmonics, for example IEC 1000-3-2, so-called pump circuits are also used which involve relatively low complexity in terms of circuitry. In principle, the rectifier is in this case coupled to the main energy store, the intermediate circuit capacitor C 6 , via an electronic pump switch. The pump nodes lying on the one hand between the diodes D 5  and D 7  and on the other hand between the diodes D 6  and D 8  are coupled to the output of an inverter (not described in more detail) via a pump network. As a result, energy is drawn from the system voltage during one half-cycle of the inverter frequency via the pump nodes and is buffer-stored in a pump network. In the subsequent half-cycle, the buffer-stored energy is fed to the intermediate circuit storage capacitor C 6  via the electronic pump switch, in this case the diodes D 8  and D 7 . Energy is thus drawn from the system with the timing of the inverter frequency. The mentioned filter elements suppress the corresponding spectral components, with the result that, finally, an almost sinusoidal system current consumption takes place. 
   Details on the pump circuit are not required for the present invention. Here, reference is made to the prior art and, in particular, to the applications DE 103 03 276.2 and DE 103 03 277.0 by the same applicant. 
   The intermediate circuit capacitor C 6  supplies to the converter which is in this case in the form of a half-bridge comprising two switching transistors V 1  and V 2 . The half-bridge transistors V 1  and V 2  produce an AC potential by corresponding clocking, in phase opposition, at their central tap, said AC potential oscillating between the two potentials of the rectifier output. This AC potential is connected to the supply branches via a lamp inductor LD 1  and, in the present case, a series circuit comprising two discharge lamps LA 1  and LA 2  and a differential current transformer TR 2  (which is explained in more detail below) via two coupling capacitors C 15 , C 16 . 
     FIG. 1  shows the fact that, in this case, not only a current can flow through the discharge plasma in the lamps LA 1  and LA 2 , but also a preheating current can flow through the upper electrode of the upper lamp LA 1  and a winding of a heating transformer TR 1  and a PTC thermistor R 1  and the lower electrode of the lower lamp LA 2 . The preheating current for the upper electrode of the lower lamp LA 2  and the lower electrode of the upper lamp LA 1  is generated by means of the heating transformer TR 1 . It can be seen in  FIG. 1  that the differential current transformer TR 2  finally determines, in its lowermost winding in  FIG. 1 , the difference between the total lamp current through the uppermost winding of the differential current transformer TR 2  and the preheating current through the central winding. In the case of only a single discharge lamp, the heating transformer TR 1  and its circuit through the inner electrodes would be dispensed with. 
   The preheating current is produced during the preheating phase, inter alia, by the value of the PTC thermistor R 1 . During the preheating phase, the value of R 1  is initially so low that a current is achieved which is predetermined by the lamp data. After the preheating phase, the value of R 1  increases such that, finally, a heating current flows which is negligible in comparison with the actual discharge current. 
   The described arrangement for preheating brings about, during the preheating phase, severe damping of a lamp resonant circuit described below and thus a reduction in the natural frequency markedly below the resonant frequency of the undamped lamp resonant circuit. During the preheating phase, an inverter frequency is used which is below the resonant frequency of the undamped lamp resonant circuit and thus ensures high heating currents and a short preheating phase. 
   The lamp resonant circuit has, in addition to the abovementioned lamp inductor LD 1 , resonant capacitors C 5  and C 9 . The resonant frequency is established by an effective capacitance comprising C 9  or the series circuit comprising C 5  and C 9 . 
   If the described lamp resonant circuit is excited after the preheating phase as a result of the damping, which is dropping off owing to the high resistance value of R 1 , and as a result of the correspondingly increased magnification factor in the vicinity of its resonant frequency, a high starting voltage is produced across the lamps LA 1  and LA 2 , and this starting voltage results in the discharge lamps being started with the aid of the preheated electrodes. Following starting, the lamp resonant circuit acts as a matching network which transforms the output impedance of the inverter to an impedance which is suitable for operation of the discharge lamps. 
   Overall, the lamp resonant circuit also acts as a pump network. If the potential across the abovementioned pump nodes is lower than the instantaneous system voltage, the pump network draws energy from the system. In the reverse case, the energy consumed is output to the intermediate circuit capacitor C 6 . A further pumping action originates from the capacitor C 8 . The capacitor C 8  continues to act as a so-called trapezoidal capacitor for relieving the switching load on the half-bridge transistors V 1  and V 2 . The pump network for the second pump branch comprises a series circuit comprising a pump inductor L 1  and a pump capacitor C 10 . 
   The half-bridge transistors V 1  and V 2 , which are designed as MOSFETs, are driven at their gates by an integrated circuit, for example of the International Rectifier IR2153 type. This control circuit also contains a high-side driver for driving the “high-side” half-bridge transistor V 1 . In this context, the diode D 9  and the capacitor C 4  are provided. 
   In addition to the driver circuits for the half-bridge transistors V 1  and V 2 , the control circuit only contains an oscillator, whose frequency can be adjusted via the connections  2  and  3  (RT and CT). This frequency corresponds to the operating frequency of the half-bridge. A frequency-determining resistor R 12  is connected between the connections  2  and  3 . A frequency-determining capacitor C 12  and, connected in series therewith, the emitter/collector path of a bipolar transistor T 3  is connected between the connection  3  and the lower supply branch acting as the reference potential. A diode D 15  is connected in parallel with the emitter/collector path in order to be able to charge and discharge C 12 . The half-bridge frequency can be adjusted using a voltage between the base connection of the bipolar transistor T 3  and the reference potential, and a manipulated variable is thus formed for a control loop. The base connection of the bipolar transistor T 3  is driven by circuit components which are illustrated further on the right in  FIG. 1 . The bipolar transistor and the control circuit as well as the associated circuitry thus form a controller. 
   The functions of the control circuit and the associated circuitry may also be realized by any desired voltage- or current-controlled oscillator circuit, which drives the converter transistors via driver circuits. 
   In the exemplary embodiment, the controller detects the lamp current as a control variable, to be precise the discharge current. Said discharge current is detected at the lowermost winding of the abovementioned differential current transformer TR 2 . A full-bridge rectifier GL 1  rectifies the current and passes it on, via a low-value measuring resistor R 21 , to the reference potential. The voltage drop across R 21  is passed to the input of a non-inverting measuring amplifier in the form of an operational amplifier U 2 -A via a low-pass filter comprising the resistors R 22  and R 32  and the capacitor C 21 , which is used for averaging purposes. Said measuring amplifier is connected in a known manner by means of the resistors R 23 -R 25  and transmits its output signal via the diode D 23  to the above-described controller input (manipulated variable node). The current control loop, which has already been referred to previously as the continuous-operation control circuit, is thus closed. The diode D 23  in this case decouples the output of the measuring amplifier U 2 -A from the voltage divider D 24 , C 20 , R 20 , D 16 , R 11  if the potential across the connection point LD 1 -D 21  is sufficiently high. According to the invention, the circuit arrangement is designed in this case such that, without a discharge current, the potential across the anode of the diode D 23  assumes the starting value. Said starting value is below a minimum value which limits the operating range of the transistor T 3  and thus the controller. Fluctuations in the potential thus have no influence on the half-bridge frequency as long as the potential remains below the minimum value. The control loop is thus not closed. The starting value brings about a half-bridge frequency which corresponds to the starting frequency. In this case, a relatively low frequency is selected via C 12  and R 12  which ensures high heating currents and short preheating phases. 
   Since the starting phase which follows on from the preheating represents a high load for the half-bridge switches V 1  and V 2  and the lamp resonant circuit LD 1 , C 5 , C 9 , a protective circuit is provided here for preventing starting voltages which are too high. However, this protective circuit at the same time also forms a voltage control circuit for adjusting the starting voltage to a suitable value. For this purpose, a varistor D 24  is used at the lamp-side connection of the lamp inductor LD 1 . Instead of a metal-oxide varistor, it is also possible in this case for a suppressor diode or a zener diode to be used, i.e. a threshold value switch. Beyond a specific threshold value, the lamp voltage is passed between two diodes D 16  via a series circuit having a capacitor C 20  and a resistor R 20 . The anode of the left-hand diode represents a second controller input. The value of the resistor R 20  influences the level of effect that the intervention, described below, has on the control loop. 
   The lamp voltage, which is tapped off via the varistor D 24 , forms a measure of the reactive energy, oscillating in the lamp resonant circuit, and of the starting voltage. If this voltage exceeds the threshold value of the varistor D 24 , the half-bridge frequency is increased and the reactive energy oscillating in the resonant circuit is thus reduced and, on the other hand, the lamp voltage is reduced. 
   A typical value for the threshold value of the varistor D 24  is, for example, 250 V. The voltage control circuit then controls the voltage such that it is above this voltage. 
   Following starting, a lamp current flows which lifts the potential across the anode of the diode D 23  to a value which is in the operating range of the bipolar transistor T 3  and thus closes the control loop of the continuous-operation control circuit (for the lamp current). 
   On the other hand, in the case of a lamp voltage, which is above the threshold value of the varistor D 24 , across the right-hand diode D 16 , which drives a tap between the resistors R 22  and R 32  at the positive input of the control amplifier U 2 -A, the potential is lifted at this input. The continuous-operation control circuit can thus be brought out of operation in accordance with the invention if the above-described situation of a new starting attempt occurs without the PTC thermistor R 1  having cooled down. 
   In such a case, only one “abnormal” glow discharge in the discharge lamps LA 1  and LA 2  would take place owing to the lack of preheating, and in this case relatively high lamp voltages would occur. This abnormal glow discharge, however, produces a notable discharge current, which is measured by means of the differential current transformer TR 2  and which brings the continuous-operation control circuit into operation. However, this would now have an influence on the half-bridge frequency and would thus finally disrupt restarting of the lamp by the frequency being moved away from the resonant frequency. 
   However, applying a (negative) component of the high lamp voltage across the components D 24 , C 20 , R 20 , D 16  to the non-inverting input of the control amplifier U 2 -A causes the continuous-operation control circuit to be blocked such that the above-described voltage control circuit remains in operation. This sets a suitable starting voltage such that the lamp can restart despite failure of the regular preheating operation. Although such a starting operation puts a strain on the electrodes, it does in the end result in the lamp operating. D 24  in this case represents a bidirectional zener diode (or suppressor diode or else a varistor) and acts as a threshold value component for decoupling purposes in different operating states. 
     FIG. 2  shows a second exemplary embodiment and differs from the first exemplary embodiment shown in  FIG. 1  as described below. For simplification purposes, reference numerals relating to elements already designated in  FIG. 1  whose function has not substantially changed are omitted. 
   As a deviation from the series connection of the two lamps LA 1  and LA 2  in  FIG. 1 , in this case the two lamps LA 1  and LA 2  are connected in parallel load circuits. No preheating transformer is therefore required; rather, direct preheating of the respective lamp electrodes takes place via the PTC thermistor R 1  for the lamp LA 1  and the PTC thermistor R 111  for the lamp LA 2 . 
   The differential current transformer TR 2 , which, however, in this case measures only the lamp current of the lamp LA 1  as a deviation from that in  FIG. 1 , acts as a device for lamp current measurement. During lamp operation, the lamp current of the lamp LA 1  thus acts as a control variable, the separate resonant circuit of the lamp LA 2  following the frequency controlled for the lamp LA 1 . However, it would also be conceivable for the controlled lamp current to be formed from components comprising (in this case) both lamp currents. 
   In this case, the separate voltage divider circuits comprising, on the one hand, C 22 , R 2 , R 9 , D 51  and, on the other hand, C 20 , R 17 , R 20 , D 50  correspond to the voltage divider circuit comprising D 24 , C 20  and R 20  in  FIG. 1 , the respectively greater potential being dominant via said circuits, to be precise via the diodes D 5  and D 13  for blocking the continuous-operation control circuit and via the diodes D 70  and D 101  having the resistor R 7  for the voltage control circuit. This is an exclusive-OR combination. 
   In this case, the coupling capacitors C 17  and C 160  are used in place of the two symmetrical coupling capacitors C 15  and C 16  in  FIG. 1 . In contrast to  FIG. 1 , here only in each case one coupling capacitor is connected to a lamp connection. However, since there is in this case a parallel circuit comprising two lamps (or more generally a parallel circuit comprising an even number of lamps), even this is a symmetrical solution which as a result does not lead to disadvantageous current loads on the storage capacitor C 6  (cf.  FIG. 1 ). 
     FIG. 3  shows a third exemplary embodiment, which differs from the first exemplary embodiment shown in  FIG. 1  as described below. In this case too, the reference numerals have been omitted. 
   Initially, in this case only one single discharge lamp LA 1  is provided, with the result that the heating transformer TR 1  in  FIG. 1  can be dispensed with. 
   In addition, there is only one pump branch, for which reason the components D 6 -D 8 , C 10 , L 1  are dispensed with. In addition, there is no differential current transformer here. Instead, the lamp current is measured in series with the coupling capacitor C 16  via a measuring resistor R 21  (to be precise the load circuit current multiplied by the factor C 16 /(C 15 +C 16 )) and passed to the base of a bipolar transistor T 4  (impedance converter), which replaces the operational amplifier U 2 -A, via a resistor R 22 . This bipolar transistor in this case acts as a control amplifier in the continuous-operation control circuit. The diodes D 7  serve the purpose of taking account of only the positive half-cycle during lamp current measurement in order to obtain a suitable potential for the control amplifier. 
   The lamp electrodes of the single lamp LA 1  are in this case preheated directly without a preheating transformer via the TISP/SIDAC D 17  and the PTC thermistor R 3 . In order to suppress the control of the load circuit current flowing when preheating and when starting the lamp LA 1 , and in order to make it possible to control the voltage via C 20 , D 24 , R 20 , D 16 , the voltage drop across the PTC thermistor R 3 , which is high in these modes of operation, is utilized in order to inject a negative current via C 17  and D 8  and thus to turn the bipolar transistor T 4  off. 
   The RC element R 22 /C 21  forms, in analogy to  FIG. 1 , the arithmetic mean of the voltage across R 21 , which is proportional to the lamp current and which is passed on to the VCO input (base T 3 ) via the emitter follower T 4 . The diode D 16  limits the negative voltage at the base of T 4  to its forward voltage, and the series circuit D 10 /D 11  dissipates the positive current half-cycle through D 17  towards the reference potential (ground) without limiting the positive voltage at the base of T 4  during operation of the lamp.