Abstract:
A semiconductor device includes first and second output stage transistors, and a first transistor, and a first constant current source, and a first specific transistor, and a second transistor, and a second constant current source and a second specific transistor. The first and second output stage transistors generate an output signal as a result of a push-pull operation, which are mutually connected in series between a first power supply and a second power supply. The first transistor has a control electrode, to which a first input signal is inputted, and is connected between the first power supply and the second power supply. The first constant current source is connected in series to the first transistor between the first power supply and the second power supply. The first specific transistor is connected in series to the first transistor and the first constant current source between the first power supply and the second power supply and is connected as current mirror to the first output stage transistor. The second transistor has a control electrode, to which a second input signal is inputted, and is connected between the first power supply and the second power supply. The second constant current source is connected in series to the second transistor between the first power supply and the second power supply. The second specific transistor is connected in series to the second transistor and the second constant current source between the first power supply and the second power supply and is connected as current mirror to the second output stage transistor.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    The present invention relates to a semiconductor device, a charge pump circuit and a PLL circuit. More particularly, the present invention relates to a semiconductor device, a charge pump circuit and a PLL circuit that can suppress a switching noise.  
           [0003]    2. Description of the Related Art  
           [0004]    As a conventional charge pump, a technique disclosed in Japanese Laid Open Patent Application (JP-A-Heisei, 11-339463) is well known. As shown in FIG. 1, when switching transistors  2 ,  3  connected to an up signal input terminal  10  and a down signal input terminal  11 , respectively, are turned on and off, the parasitic capacitances of the respective transistors  2 ,  3  cause the voltages of drain terminals  12 ,  13  of constant current transistors  1 ,  4  to be different from each other.  
           [0005]    For this reason, the current values are different when on-signals are inputted to the up signal input terminal  10  and the down signal input terminal  11 , respectively. This difference brings about a generation of an error signal.  
           [0006]    In order to solve the above-mentioned problem, conventionally, as shown in FIG. 2, a first condenser  6  is mounted between the drain of the constant current transistor  1  and a positive power supply terminal VDD, a second condenser  7  is mounted between the drain of the constant current transistor  1  and a negative power supply terminal VSS, a third condenser  8  is mounted between the drain of the constant current transistor  4  and the positive power supply terminal VDD, and a fourth condenser  9  is mounted between the drain of the constant current transistor  4  and the negative power supply terminal VSS.  
           [0007]    The first and second condensers  6 ,  7  are used to keep the drain voltage and the source voltage of the switching transistor  2  substantially equal to each other. The third and fourth condensers  8 ,  9  are used to keep the drain voltage and the source voltage of the switching transistor  3  substantially equal to each other. Here, the condenser capacitances of the first to fourth condensers  6  to  9  are designed to be very larger than the parasitic capacitances parasitic on the drain terminals of the drain terminals  12 ,  13  of the constant current transistors  1 ,  4 .  
           [0008]    Accordingly, it is possible to reduce the voltage variations in the drain terminals  12 ,  13  when the on-signals are inputted to the up signal input terminal  10  and the down signal input terminal  11 , respectively, and thereby possible to suppress the error current.  
           [0009]    According to the above-mentioned configuration, a relatively large capacitance, such as several p to several ten pF, is required to thereby result in a problem that an area for it is divided.  
           [0010]    Also, the measure for the parasitic capacitances of the switching transistors  2 ,  3  that cause the switching noise is not carried out, which results in a problem that the switching noise is not suppressed. In the above-mentioned configuration, a switching voltage is 0 to VDD, and this is a very high value. Thus, the high voltage charged in the parasitic capacitance leads to the switching noise.  
           [0011]    By the way, Japanese Laid Open Patent Application (JP-A-2000-49596) discloses the following charge pump circuit. This is a charge pump circuit used in a PLL circuit. This turns back a current generated from a constant current source by using a current mirror circuit, via an analog switch that is always turned on, and in response to an UP signal sent from a frequency/phase comparator, sends/stops a constant current, which is turned ON/OFF by a switching circuit containing an analog switch and copied by a current mirror, towards a loop filter at a latter stage, and similarly in response to a DOWN signal, sends/stops a constant current, which is turned ON/OFF by a switching circuit containing an analog switch and copied by a current mirror, towards a loop filter at a latter stage.  
           [0012]    A charge pump is desired for suppressing a switching noise.  
           [0013]    A charge pump is desired for suppressing a switching noise without any necessity of a wide area.  
         SUMMARY OF THE INVENTION  
         [0014]    The present invention is accomplished in view of the above mentioned problems. Therefore, an object of the present invention is to provide a semiconductor device, a charge pump and a PLL circuit which can suppress a switching noise.  
           [0015]    Another object of the present invention is to provide a semiconductor device, a charge pump and a PLL circuit which can suppress a switching noise without any necessity of a wide area.  
           [0016]    Still another object of the present invention is to provide a semiconductor device, a charge pump and a PLL circuit which can suppress a switching noise and compensate for an error current.  
           [0017]    Still another object of the present invention is to provide a semiconductor device, a charge pump and a PLL circuit which can suppress a switching noise without any necessity of a wide area and compensate for an error current.  
           [0018]    In order to achieve an aspect of the present invention, a semiconductor device, includes: first and second output stage transistors generating an output signal as a result of a push-pull operation, which are mutually connected in series between a first power supply and a second power supply; a first transistor that has a control electrode, to which a first input signal is inputted, and is connected between the first power supply and the second power supply; a first constant current source connected in series to the first transistor between the first power supply and the second power supply; a first specific transistor which is connected in series to the first transistor and the first constant current source between the first power supply and the second power supply and connected as current mirror to the first output stage transistor; a second transistor that has a control electrode, to which a second input signal is inputted, and is connected between the first power supply and the second power supply; a second constant current source that is connected in series to the second transistor between the first power supply and the second power supply; and a second specific transistor which is connected in series to the second transistor and the second constant current source between the first power supply and the second power supply and connected as current mirror to the second output stage transistor.  
           [0019]    In this case, the semiconductor device, further includes: a current error compensation circuit compensating for errors of currents respectively flowing through the first and second output stage transistors at the time of the push-pull operation, in accordance with the output signal and a referential signal.  
           [0020]    Also in this case, the first transistor and the second transistor are MOS-type transistors.  
           [0021]    In order to achieve another aspect of the present invention, a charge pump circuit that is used in a PLL (Phase-Locked Loop) circuit, and generates an output signal in response to an up instruction signal and a down instruction signal sent from a phase comparator to drive a VCO (Voltage-Controlled Oscillator) in accordance with the output signal, includes: first and second output stage transistors generating the output signal as a result of a push-pull operation, which are mutually connected in series between a first power supply and a second power supply; a first transistor that has a control electrode, to which the up instruction signal is inputted, and is connected between the first power supply and the second power supply; a first constant current source that is connected in series to the first transistor between the first power supply and the second power supply; a first specific transistor which is connected in series to the first transistor and the first constant current source between the first power supply and the second power supply and connected as current mirror to the first output stage transistor; a second transistor that has a control electrode, to which an inversion signal of the down instruction signal is inputted, and is connected between the first power supply and the second power supply; a second constant current source that is connected in series to the second transistor between the first power supply and the second power supply; and a second specific transistor which is connected in series to the second transistor and the second constant current source between the first power supply and the second power supply and connected as current mirror to the second output stage transistor.  
           [0022]    In this case, the charge pump circuit, further includes: a current error compensation circuit compensating for errors of currents respectively flowing through the first and second output stage transistors at the time of the push-pull operation, in accordance with the output signal and a referential signal.  
           [0023]    Also in this case, the first transistor and the second transistor are MOS-type transistors.  
           [0024]    In order to achieve still another aspect of the present invention, a PLL (Phase-Locked Loop) circuit, includes: a phase comparator; a VCO (Voltage Controlled Oscillator); and a charge pump circuit generating an output signal in response to an up instruction signal and a down instruction signal sent from the phase comparator to drive the VCO based on the output signal, and wherein the charge pump circuit includes: first and second output stage transistors generating the output signal as a result of a push-pull operation, which are mutually connected in series between a first power supply and a second power supply; a first transistor that has a control electrode, to which the up instruction signal is inputted, and is connected between the first power supply and the second power supply; a first constant current source that is connected in series to the first transistor between the first power supply and the second power supply; a first specific transistor which is connected in series to the first transistor and the first constant current source between the first power supply and the second power supply and connected as current mirror to the first output stage transistor; a second transistor that has a control electrode, to which an inversion signal of the down instruction signal is inputted, and is connected between the first power supply and the second power supply; a second constant current source that is connected in series to the second transistor between the first power supply and the second power supply; and a second specific transistor which is connected in series to the second transistor and the second constant current source between the first power supply and the second power supply and connected as current mirror to the second output stage transistor.  
           [0025]    In this case, the PLL circuit, further includes: a current error compensation circuit compensating for errors of currents respectively flowing through the first and second output stage transistors at the time of the push-pull operation, in accordance with the output signal and a referential signal.  
           [0026]    Also in this case, the first transistor and the second transistor are MOS-type transistors.  
           [0027]    In order to achieve yet still another aspect of the present invention, a semiconductor device, includes: first and second output stage transistors generating a first output signal as a result of a push-pull operation, which are mutually connected in series between a first power supply and a second power supply; third and fourth output stage transistors generating a second output signal as a result of a push-pull operation, which are mutually connected in series between the first power supply and the second power supply; first and second differential transistor pairs that conductive types are opposite to each other and have control electrodes connected to first and second input terminals, respectively; first and second constant current sources connected to the first and second differential transistor pairs, respectively; a first current mirror circuit connected between the first differential transistor pairs and the first power supply; a second current mirror circuit connected between the second differential transistor pairs and the second power supply; third and fourth differential transistor pairs that conductive types are opposite to each other and have control electrodes connected to third and fourth input terminals, respectively; third and fourth constant current sources connected to the third and fourth differential transistor pairs, respectively; a third current mirror circuit connected between the third differential transistor pairs and the first power supply; and a fourth current mirror circuit connected between the fourth differential transistor pairs and the second power supply, and wherein the first output stage transistor is included in the first current mirror circuit, and wherein the second output stage transistor is included in the fourth current mirror circuit, and wherein the third output stage transistor is included in the third current mirror circuit and wherein the fourth output stage transistor is included in the second current mirror circuit.  
           [0028]    In this case, the semiconductor device, further includes: a current error compensation circuit compensating for errors of currents respectively flowing through the first and second output stage transistors at the time of the push-pull operation, in accordance with the output signal and a referential signal.  
           [0029]    In order to achieve another aspect of the present invention, a charge pump circuit that is used in a PLL (Phase-Locked Loop) circuit, and generates a first output signal and a second output signal in which the first output signal is inverted, in response to an up instruction signal and a down instruction signal sent from a phase comparator to drive a VCO (Voltage-Controlled Oscillator) in accordance with the first and second output signals, includes: first and second output stage transistors generating the first output signal as a result of a push-pull operation, which are mutually connected in series between a first power supply and a second power supply; third and fourth output stage transistors generating the second output signal as a result of a push-pull operation, which are mutually connected in series between the first power supply and the second power supply; first and second differential transistor pairs that conductive types are opposite to each other and have control electrodes connected to first and second input terminals, respectively, to which the up instruction signal and an up instruction inversion signal where the up instruction signal is inverted are sent; first and second constant current sources connected to the first and second differential transistor pairs, respectively; a first current mirror circuit connected between the first differential transistor pairs and the first power supply; a second current mirror circuit connected between the second differential transistor pairs and the second power supply; third and fourth differential transistor pairs that conductive types are opposite to each other and have a control electrode connected to third and fourth input terminals, respectively, to which the down instruction signal and a down instruction inversion signal where the down instruction signal is inverted are sent; third and fourth constant current sources connected to the third and fourth differential transistor pairs, respectively; a third current mirror circuit connected between the third differential transistor pairs and the first power supply; and a fourth current mirror circuit connected between the fourth differential transistor pairs and the second power supply, and wherein the first output stage transistor is included in the first current mirror circuit, and wherein the second output stage transistor is included in the fourth current mirror circuit, and wherein the third output stage transistor is included in the third current mirror circuit and wherein the fourth output stage transistor is included in the second current mirror circuit.  
           [0030]    In this case, the charge pump circuit, further includes: a current error compensation circuit compensating for errors of currents flowing through the first to fourth mirror circuits, respectively.  
           [0031]    Also in this case, the charge pump circuit, further includes: fifth and sixth constant current sources connected parallel to the first and third constant current sources, respectively; and wherein the current error compensation circuit generates a control signal corresponding to a difference between a set signal and a signal indicative of an average value between the first and second output signals, and wherein the fifth and sixth constant current sources change values of currents to be sent to the first and third differential transistor pairs, in response to the control signal.  
           [0032]    In order to achieve still another aspect of the present invention, a PLL (Phase-Locked Loop) circuit, includes: a phase comparator; a VCO (Voltage-Controlled Oscillator); and a charge pump circuit generating a first output signal and a second output signal in which the first output signal is inverted, in response to an up instruction signal and a down instruction signal sent from the phase comparator to drive the VCO in accordance with the first and second output signals, and wherein the charge pump circuit includes: first and second output stage transistors generating the first output signal as a result of a push-pull operation, which are mutually connected in series between a first power supply and a second power supply; third and fourth output stage transistors generating the second output signal as a result of a push-pull operation, which are mutually connected in series between the first power supply and the second power supply; first and second differential transistor pairs that conductive types are opposite to each other and have control electrodes connected to first and second input terminals, respectively, to which the up instruction signal and an up instruction inversion signal where the up instruction signal is inverted are sent; first and second constant current sources connected to the first and second differential transistor pairs, respectively; a first current mirror circuit connected between the first differential transistor pairs and the first power supply; a second current mirror circuit connected between the second differential transistor pairs and the second power supply; third and fourth differential transistor pairs that conductive types are opposite to each other and have control electrodes connected to third and fourth input terminals, respectively, to which the down instruction signal and a down instruction inversion signal where the down instruction signal is inverted are sent; third and fourth constant current sources connected to the third and fourth differential transistor pairs, respectively; a third current mirror circuit connected between the third differential transistor pairs and the first power supply; and a fourth current mirror circuit connected between the fourth differential transistor pairs and the second power supply, and wherein the first output stage transistor is included in the first current mirror circuit, and wherein the second output stage transistor is included in the fourth current mirror circuit, and wherein the third output stage transistor is included in the third current mirror circuit and wherein the fourth output stage transistor is included in the second current mirror circuit.  
           [0033]    In order to achieve yet still another aspect of the present invention, the PLL circuit, further includes: a current error compensation circuit compensating for errors of currents flowing through the first to fourth mirror circuits, respectively.  
           [0034]    In the present invention, the charge pump circuit is constituted by using the current mirror in order to suppress the switching noise. Moreover, the error current compensation circuit is used in order to compensate for the error current in the charged case.  
           [0035]    The UP, UPB, DOWN and DOWNB signals are inputted from the phase comparator to the differential circuits. The load of the differential circuit is constituted by the current mirror circuit. So, the current flowing through the load is outputted to the filter of the PLL.  
           [0036]    Also, the differential circuit has the current error compensation terminal for compensating for the current errors on the UP side and the DOWN side. The operation for compensating for the current error compares a middle value of the capacitance terminal voltages of the respective filters with a reference voltage (ref), and its compared result is fed back to the charge pump (Common Mode Feed Back). 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0037]    [0037]FIG. 1 is a circuit diagram of a conventional charge pump circuit;  
         [0038]    [0038]FIG. 2 is a circuit diagram showing another conventional charge pump circuit;  
         [0039]    [0039]FIG. 3 is a circuit diagram showing a conventional typical PLL circuit;  
         [0040]    [0040]FIG. 4 is a circuit diagram showing a charge pump circuit of a first embodiment of the present invention;  
         [0041]    [0041]FIG. 5 is a circuit diagram showing a current error compensation circuit connected to the charge pump circuit of the first embodiment of the present invention; and  
         [0042]    [0042]FIG. 6 is a circuit diagram showing in detail the current error compensation circuit of the first embodiment of the present invention. 
     
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0043]    Embodiments of the present invention will be described below in detail with reference to the attached drawings.  
         [0044]    A charge pump circuit in this embodiment is a charge pump circuit composed of MOS-type transistors for driving a variable frequency oscillator via a loop filter, in response to a down instruction (DOWN) signal and an up instruction (UP) signal sent from a frequency phase comparator of a PLL (Phase-Locked Loop) circuit locked at a phase of an input signal.  
         [0045]    At first, a conventionally typical PLL circuit is described with reference to FIG. 3.  
         [0046]    A PLL circuit  100  contains a first or second loop filter  120  having resistors and condensers, a VCO (Voltage Controlled Oscillator)  130 , a division circuit  140 , a frequency phase detection circuit  150 , a charge pump  110  and the like.  
         [0047]    The loop filter  120  functions as a low pass filter and generates a terminal voltage of a condenser C 2  as a control voltage.  
         [0048]    The division circuit  140  divides an oscillation signal of the VCO  130  on the basis of a division ratio, and generates the divided signal as an output signal.  
         [0049]    The frequency phase detection circuit  150  detects the errors in frequencies and phases between an input signal and a signal in which the oscillation signal of the VCO  130  is divided by the division circuit  140 , and generates an up instruction (UP) signal and a down instruction (DOWN) signal, on the basis of the errors.  
         [0050]    The charge pump  110  implants a certain current from a power supply VDD into the loop filter  120 , in response to the up instruction (UP) signal, and accumulates charges in condensers C 1 , C 2 . Also, the charge pump  110  discharges the certain current from the loop filter  120 , in response to the down instruction (DOWN) signal, and discharges the charges accumulated in the condensers C 1 , C 2 , and implants into the charge pump  110 .  
         [0051]    The above-mentioned series of operations enables the components of the phases and the frequencies in the input signal and the output signal to be coincident with each other, in the PLL circuit  100  (Synchronous State).  
         [0052]    A charge pump circuit in this embodiment will be described below with reference to FIGS. 4 and 5. The charge pump circuit in this embodiment is applied to a PLL circuit for a high speed communication.  
         [0053]    As shown in FIG. 4, the charge pump circuit is provided with an up stage K 1  and a down stage K 2 .  
         [0054]    As shown in FIG. 5, output signals outputted from an output terminal C and an output terminal (an inversion terminal of the output terminal C) CB of the charge pump circuit are outputted through a PLL filter  50  to the VCO. By the way, the PLL filter  50  is a low pass filter composed of a resistor R and a condenser CO.  
         [0055]    At first, the up stage K 1  of the charge pump circuit is described with reference to FIG. 4.  
         [0056]    In the up stage K 1 , a differential amplifier composed of N-channel transistors MN 12 , MN 13  is connected parallel to a differential amplifier composed of P-channel transistors MP 13 , MP 14 .  
         [0057]    The N-channel transistors MN 12 , MN 13  constitute the differential amplifier. Sources of the N-channel transistors MN 12 , MN 13  are connected to each other. A gate of the N-channel transistor MN 12  is connected to an input terminal (UP) A 1 . The up instruction (UP) signal is inputted to the input terminal A 1 . A gate of the N-channel transistor MN 13  is connected to an input terminal (UPB) A 2 . An inversion signal of the up instruction (UP) signal is inputted to the input terminal A 2 .  
         [0058]    In an N-channel transistor MN 10  for a constant current source, its source is connected to a low potential side power supply AVSS, and its drain is connected to the commonly connected sources of the N-channel transistors MN 12 , MN 13 . The N-channel transistor MN 10  for the constant current source is a current suck type, and a current of 10 μA flows through it.  
         [0059]    P-channel transistors MP 10 , MP 1  constitute a current mirror circuit functioning as the positive load of the differential amplifiers MN 12 , MN 13 . In each of the P-channel transistors MP 10 , MP 1 , its source is connected to a high potential side power supply AVDD. A gate of the P-channel transistor MP 1  is connected to the drain of the N-channel transistor MN 12 , and its drain is connected to an output terminal C. A drain of the P-channel transistor MP 10  is connected to the drain of the N-channel transistor MN 12 . The gate and the drain of the P-channel transistor MP 10  are connected to each other.  
         [0060]    In the P-channel transistor MP 11 , its drain is connected to the drain of the N-channel transistor MN 13 , and its source is connected to the high potential side power supply AVDD. In the P-channel transistor MP 11 , the property equal to that of the P-channel transistor MP 10  is used in order to make the loads of the differential amplifiers MN 12 , MN 13  equal to each other.  
         [0061]    In the N-channel transistor MN 11  for the constant current source, its drain is connected to the sources of the N-channel transistors MN 13 , NM 12 , and its source is connected to the low potential side power supply AVSS. The N-channel transistor NM 11  for the constant current source and an N-channel transistor MN 26  of FIG. 6 constitute a current mirror circuit.  
         [0062]    The P-channel transistors MP 13 , MP 14  constitute the differential amplifier. The sources of the P-channel transistors MP 13 , MP 14  are connected to each other. The gate of the P-channel transistor MP 13  is connected to the input terminal A 1 . The gate of the P-channel transistor MP 14  is connected to the input terminal A 2 .  
         [0063]    In the P-channel transistor MP 12  for the constant current source, its source is connected to the high potential side power supply AVDD, and its drain is connected to the commonly connected sources of the P-channel transistors MP 13 , MP 14 . A current of 10 μA flows through the P-channel transistor MP 12  for the constant current source.  
         [0064]    N-channel transistors MN 15 , MN 2  constitute a current mirror circuit functioning as the positive load of the differential amplifiers MP 13 , MP 14 . In each of the N-channel transistors MN 15 , MN 2 , its source is connected to the low potential side power supply AVSS. A gate of the N-channel transistor MN 2  is connected to the drain of the P-channel transistor MP 14 , and its drain is connected to an output terminal CB. A drain of the N-channel transistor MN 15  is connected to the drain of the P-channel transistor MP 14 . The gate and the drain of the N-channel transistor MN 15  are connected to each other.  
         [0065]    In the N-channel transistor MN 14 , its drain is connected to the drain of the P-channel transistor MP 13 , and its source is connected to the low potential side power supply AVSS. In the N-channel transistor MN 14 , the property equal to that of the N-channel transistor MN 15  is used in order to make the loads of the differential amplifiers MP 13 , MP 14  equal to each other.  
         [0066]    The down stage K 2  of the charge pump circuit will be described below.  
         [0067]    In the down stage K 2 , a differential amplifier composed of N-channel transistors MN 18 , MN 19  is connected parallel to a differential amplifier composed of P-channel transistors MP 18 , MP 19 .  
         [0068]    The N-channel transistors MN 18 , MN 19  constitute the differential amplifier. Sources of the N-channel transistors MN 18 , MN 19  are connected to each other. A gate of the N-channel transistor MN 18  is connected to an input terminal (DOWN) A 3 . The down instruction (DOWN) signal is inputted to the input terminal A 3 . A gate of the N-channel transistor MN 19  is connected to an input terminal (DOWNB) A 4 . An inversion signal of the down instruction (DOWN) signal is inputted to the input terminal A 4 .  
         [0069]    In an N-channel transistor MN 16  for a constant current source, its source is connected to a low potential side power supply AVSS, and its drain is connected to the commonly connected sources of the N-channel transistors MN 18 , MN 19 . The N-channel transistor MN 16  for the constant current source is the current suck type, and a current of 10 μA flows through it.  
         [0070]    P-channel transistors MP 15 , MP 2  constitute a current mirror circuit functioning as the positive load of the differential amplifiers MN 18 , MN 19 . In each of the P-channel transistors MP 15 , MP 2 , its source is connected to a high potential side power supply AVDD. A gate of the P-channel transistor MP 2  is connected to the drain of the N-channel transistor MN 18 , and its drain is connected to an output terminal CB. A drain of the P-channel transistor MP 15  is connected to the drain of the N-channel transistor MN 18 . The gate and the drain of the P-channel transistor MP 15  are connected to each other.  
         [0071]    In the P-channel transistor MP 16 , its source is connected to the high potential side power supply AVDD, and its drain is connected to the drain of the N-channel transistor MN 19 . In the P-channel transistor MP 16 , the property equal to that of the P-channel transistor MP 15  is used in order to make the loads of the differential amplifiers MN 18 , MN 19  equal to each other.  
         [0072]    In the N-channel transistor MN 17  for the constant current source, its source is connected to the low potential side power supply AVSS, and its drain is connected to the sources of the N-channel transistors MN 18 , NM 19 . The N-channel transistor MN 17  for the constant current source and the N-channel transistor MN 26  of FIG. 6 constitute a current mirror circuit.  
         [0073]    The P-channel transistors MP 18 , MP 19  constitute the differential amplifier. The sources of the P-channel transistors MP 18 , MP 19  are connected to each other. The gate of the P-channel transistor MP 18  is connected to the input terminal A 3 . The gate of the P-channel transistor MP 19  is connected to the input terminal A 4 .  
         [0074]    In the P-channel transistor MP 17  for the constant current source, its source is connected to the high potential side power supply AVDD, and its drain is connected to the commonly connected sources of the P-channel transistors MP 18 , MP 19 . A current of 10 μA flows through the P-channel transistor MP 17  for the constant current source.  
         [0075]    N-channel transistors MN 21 , MN 1  constitute a current mirror circuit functioning as the positive load of the differential amplifiers MP 18 , MP 19 . In each of the N-channel transistors MN 21 , MN 1 , its source is connected to the low potential side power supply AVSS. A gate of the N-channel transistor MN 1  is connected to the drain of the P-channel transistor MP 19 , and its drain is connected to an output terminal C. A drain of the N-channel transistor MN 21  is connected to the drain of the P-channel transistor MP 19 . The gate and the drain of the N-channel transistor MN 21  are connected to each other.  
         [0076]    In the N-channel transistor MN 20 , its source is connected to the low potential side power supply AVSS, and its drain is connected to the drain of the P-channel transistor MP 18 . In the N-channel transistor MN 20 , the property equal to that of the N-channel transistor MN 21  is used in order to make the loads of the differential amplifiers MP 18 , MP 19  equal to each other.  
         [0077]    By the way, the current flows from an input terminal A 0  to the charge pump circuit. The circuit (group) between the input terminal A 0 , the N-channel transistor MN 10  for the constant current source, the P-channel transistor MP 12  for the constant current source, the N-channel transistor MN 16  for the constant current source and the P-channel transistor MP 17  for the constant current source enables the values of the currents (the values of the constant currents) flowing through the transistors MN 10 , MP 12 , MN 16  and MP 17  for the respective constant current sources to set at 10 μA.  
         [0078]    The operations of the present invention will be described below.  
         [0079]    The current mirror circuits MP 10 , MP 1  are connected instead of the load resistors of the differential amplifiers MN 12 , MN 13 . The action of the current mirror makes a drain current IdP 1  of the transistor MP 1  equal to a drain current IdP 10  of the transistor MP 10 .  
         [0080]    The current mirror circuits MN 15 , MN 2  are connected instead of the load resistors of the differential amplifiers MP 13 , MP 14 . The action of the current mirror makes a drain current IdN 2  of the transistor MN 2  equal to a drain current IdN 15  of the transistor MN 15 .  
         [0081]    The current mirror circuits MP 15 , MP 2  are connected instead of the load resistors of the differential amplifiers MN 18 , MN 19 . The action of the current mirror makes a drain current IdP 2  of the transistor MP 2  equal to a drain current IdP 15  of the transistor MP 15 .  
         [0082]    The current mirror circuits MN 21 , MN 1  are connected instead of the load resistors of the differential amplifiers MP 18 , MP 19 . The action of the current mirror makes a drain current IdN 1  of the transistor MN 1  equal to a drain current IdN 21  of the transistor MN 21 .  
         [0083]    At first, a case when a voltage higher than that of the input terminal (UPB) A 2  is applied to the input terminal (UP) A 1  is described. At this time, a voltage lower than that of the input terminal (DOWNB) is applied to the input terminal (DOWN) A 3 .  
         [0084]    When a signal of a high voltage is inputted to the input terminal (UP) A 1 , a gate voltage of the N-channel transistor MN 12  is made higher. Thus, the currents of 10 μA flow from the high potential side power supply AVDD into the P-channel transistor MP 10 , the N-channel transistor MN 12  and the N-channel transistor MN 10  for the constant current source. At this time, the action of the current mirror causes the drain current IdP 1  of the P-channel transistor MP 1  to be 10 μA equal to the drain current Id 10  of the P-channel transistor MP 10 .  
         [0085]    When a signal of a high voltage is inputted to the input terminal (UP) A 1 , a signal of a low voltage that is its inversion signal is inputted to the input terminal (UPB) A 2 . Thus, a gate voltage of the N-channel transistor MN 13  is made lower. Hence, the currents do not substantially flow from the high potential side power supply AVDD into the P-channel transistor MP 11  and the N-channel transistor MN 13 .  
         [0086]    When a signal of a high voltage is inputted to the input terminal (UP) A 1 , a gate voltage of the P-channel transistor MP 13  is made higher. Thus, the currents do not substantially flow into the P-channel transistor MP 13  and the N-channel transistor MN 14 .  
         [0087]    When a signal of a high voltage is inputted to the input terminal (UP) A 1 , the signal of the low voltage that is its inversion signal is inputted to the input terminal (UPB) A 2 . Thus, a gate voltage of the P-channel transistor MP 14  is made lower. Hence, the currents of 10 μA flow from the high potential side power supply AVDD into the P-channel transistor MP 12  for the constant current source, the P-channel transistor MP 14  and the N-channel transistor MN 15 . At this time, the current mirror causes a drain current IdN 2  of the N-channel transistor MN 12  to be 10 μA equal to a drain current IdN 15  of the N-channel transistor MN 15 .  
         [0088]    When a signal of a low voltage is inputted to the input terminal (DOWN) A 3 , a gate voltage of the N-channel transistor MN 18  is made lower. Thus, the currents do not substantially flow from the high potential side power supply AVDD into the P-channel transistor MP 15 , the N-channel transistor MN 18  and the N-channel transistor MN 16  for the constant current source. At this time, the current mirror causes a drain current IdP 2  of the P-channel transistor MP 2  to be substantially 0 μA equal to a drain current IdP 15  of the P-channel transistor MP 15 .  
         [0089]    When a signal of a low voltage is inputted to the input terminal (DOWN) A 3 , a signal of a high voltage that is its inversion signal is inputted to the input terminal (DOWNB) A 4 . Thus, a gate voltage of the N-channel transistor MN 19  is made higher. Hence, the currents of 10 μA flow from the high potential side power supply AVDD into the P-channel transistor MP 16 , the N-channel transistor MN 19  and the N-channel transistor MN 16  for the constant current source.  
         [0090]    When a signal of a low voltage is inputted to the input terminal (DOWN) A 3 , a gate voltage of the P-channel transistor MP 18  is made lower. Thus, the currents of 10 μA flow from the high potential side power supply AVDD into the P-channel transistor MP 17  for the constant current source, the P-channel transistor MP 18  and the N-channel transistor MN 20 .  
         [0091]    When a signal of a low voltage is inputted to the input terminal (DOWN) A 3 , a signal of a high voltage that is its inversion signal is inputted to the input terminal (DOWNB) A 4 . Thus, a gate voltage of the P-channel transistor MP 19  is made higher. Hence, the currents do not substantially flow into the P-channel transistor MP 19  and the N-channel transistor MN 21 . At this time, the current mirror causes a drain current IdN 1  of the N-channel transistor MN 1  to be substantially 0 μA equal to a drain current Id 21  of the N-channel transistor MN 21 .  
         [0092]    From the above-mentioned explanations, the following operations are carried out when the voltage higher than that of the input terminal (UPB) A 2  is applied to the input terminal (UP) A 1  and the voltage lower than that of the input terminal (DOWNB) A 4  is applied to the input terminal (DOWN) A 3 .  
         [0093]    The drain current IdP 1  of the P-channel transistor MP 1  of 10 μA flows into the output terminal C from the high potential side power supply AVDD. At this time, the current flowing into the low potential side power supply AVSS from the output terminal C through the N-channel transistor MN 1  is at the cut state (the drain current IdN 1  is very small). Thus, the current flowing into the P-channel transistor MP 1  from the high potential side power supply AVDD flows into the output terminal C. Hence, the potential of the output terminal C is increased.  
         [0094]    Also, the drain current IdN 2  of the N-channel transistor MN 2  of 10 μA flows into the low potential side power supply AVSS from the output terminal CB through the N-channel transistor MN 2 . At this time, the current flowing into the output terminal CB from the high potential side power supply AVDD through the P-channel transistor MP 2  is cut off (the drain current IdP 2  is very small). Thus, the fact that the current of 10 μA flows into the low potential side power supply AVSS from the output terminal CB through the N-channel transistor MN 2  causes the potential of the output terminal CB to be decreased.  
         [0095]    The case when the voltage lower than that of the input terminal (UPB) A 2  is applied to the input terminal (UP) A 1  will be described below. At this time, the voltage higher than that of the input terminal (DOWNB) A 4  is applied to the input terminal (DOWN) A 3 .  
         [0096]    When a signal of a low voltage is inputted to the input terminal (UP) A 1 , a gate voltage of the N-channel transistor MN 12  is made lower. Thus, the currents do not substantially flow from the high potential side power supply AVDD into the P-channel transistor MP 10 , the N-channel transistor MN 12  and the N-channel transistor MN 10  for the constant current source. At this time, the current mirror causes a drain current IdP 1  of the P-channel transistor MP 1  to be substantially 0 μA equal to a drain current Id 10  of the P-channel transistor MP 10 .  
         [0097]    When a signal of a low voltage is inputted to the input terminal (UP) A 1 , a signal of a high voltage that is its inversion signal is inputted to the input terminal (UPB) A 2 . Thus, a gate voltage of the N-channel transistor MN 13  is made higher. Hence, the currents of 10 μA flow from the high potential side power supply AVDD into the P-channel transistor MP 11 , the N-channel transistor MN 13  and the N-channel transistor MN 10  for the constant current source.  
         [0098]    When a signal of a low voltage is inputted to the input terminal (UP) A 1 , a gate voltage of the P-channel transistor MN 13  is made lower. Thus, the current of 10 μA flows from the high potential side power supply AVDD into the P-channel transistor MP 12  for the constant current source, the P-channel transistor MP 13  and the N-channel transistor MN 14 .  
         [0099]    When a signal of a low voltage is inputted to the input terminal (UP) A 1 , a signal of a high voltage that is its inversion signal is inputted to the input terminal (UPB) A 2 . Thus, a gate voltage of the P-channel transistor MN 14  is made higher. Hence, the currents do not substantially flow into the P-channel transistor MP 14  and the N-channel transistor MN 15 . At this time, the current mirror causes a drain current IdN 2  of the N-channel transistor MN 2  to be substantially 0 μA equal to a drain current IdN 15  of the N-channel transistor MN 15 .  
         [0100]    When a signal of a high voltage is inputted to the input terminal (DOWN) A 3 , a gate voltage of the N-channel transistor MN 18  is made higher. Thus, the currents of 10 μA flow from the high potential side power supply AVDD into the P-channel transistor MP 15 , the N-channel transistor MN 18  and the N-channel transistor MN 16  for the constant current source. At this time, the current mirror causes a drain current IdP 2  of the P-channel transistor MP 2  to be 10 μA equal to a drain current IdP 15  of the P-channel transistor MP 15 .  
         [0101]    When a signal of a high voltage is inputted to the input terminal (DOWN) A 3 , a signal of a low voltage that is its inversion signal is inputted to the input terminal (DOWNB) A 4 . Thus, a gate voltage of the N-channel transistor MN 19  is made lower. Hence, the currents do not substantially flow from the high potential side power supply AVDD into the P-channel transistor MP 16  and the N-channel transistor MN 19 .  
         [0102]    When a signal of a high voltage is inputted to the input terminal (DOWN) A 3 , a gate voltage of the P 20  channel transistor MP 18  is made higher. Thus, the currents do not substantially flow into the P-channel transistor MP 18  and the N-channel transistor MN 20 .  
         [0103]    When a signal of a high voltage is inputted to the input terminal (DOWN) A 3 , a signal of a low voltage that is its inversion signal is inputted to the input terminal (DOWNB) A 4 . Thus, a gate voltage of the P-channel transistor MN 19  is made lower. Hence, the currents of 10 μA flow from the high potential side power supply AVDD into the P-channel transistor MP 17  for the constant current source, the P-channel transistor MP 19  and the N-channel transistors MN 21 . At this time, the current mirror causes a drain current IdN 1  of the N-channel transistor MN 1  to be 10 μA equal to a drain current Id 21  of the N-channel transistor MN 21 .  
         [0104]    From the above-mentioned explanations, the following operations are carried out when the voltage lower than that of the input terminal (UPB) A 2  is applied to the input terminal (UP) A 1  and the voltage higher than that of the input terminal (DOWNB) A 4  is applied to the input terminal (DOWN) A 3 .  
         [0105]    The drain current IdP 2  of the P-channel transistor MP 2  of 10 μA flows into the output terminal CB from the high potential side power supply AVDD. At this time, the current flowing into the low potential side power supply AVSS from the output terminal CB through the N-channel transistor MN 2  is at the cut state (the drain current IdN 2  is very small). Thus, the current flowing into the P-channel transistor MP 2  from the high potential side power supply AVDD flows into the output terminal CB. Hence, the potential of the output terminal CB is increased.  
         [0106]    Also, the drain current IdN 1  of the N-channel transistor MN 1  of 10 μA flows into the low potential side power supply AVSS from the output terminal C through the N-channel transistor MN 1 . At this time, the current flowing into the output terminal C from the high potential side power supply AVDD through the P-channel transistor MP 1  is cut off (the drain current IdP 1  is very small). Thus, the fact that the current of 10 μA flows into the low potential side power supply AVSS from the output terminal C through the N-channel transistor MN 1  causes the potential of the output terminal C to be decreased.  
         [0107]    Conventionally, the switching voltages when the UP signal and the DOWN signal are inputted from the phase comparator to the gate of the switching transistor are high such as 0 to VDD. Thus, the high voltage charged in the parasitic condenser brings about the switching noise.  
         [0108]    On the contrary, in this embodiment, the load of the differential amplifier is constituted by the current mirror circuit. Thus, the change in the switching voltage between 0 and VDD can be converted into the change of the small current (0 to 10 μA in this embodiment). Its small current is outputted to the PLL filter  50  from the output terminals C and CB. At this time, the changes in the gate potentials of the transistors (MP 1 , MP 2 , MN 1  and MN 2 ) constituting the current mirror circuit are 500 mV (in a case of a 3.3 power supply). Thus, the amplitude of the gate potential is equal to or less than ⅙ that of the conventional circuit. Hence, the switching noise can be also reduced to the degree similar to that of the conventional circuit.  
         [0109]    A current error compensation circuit (Common Mode Feed Back Circuit) will be described below with reference to FIGS.  4  to  6 .  
         [0110]    The current error compensation circuit  60  is connected to the above-mentioned charge pump circuit (FIG. 4).  
         [0111]    As mentioned above, the output stage for sending the output signal to the output terminal C is constituted by the push-pull transistors composed of the P-channel transistor MP 1  and the N-channel transistor MN 1 . Similarly, the output stage for sending the output signal to the output terminal CB is constituted by the push-pull transistors composed of the P-channel transistor MP 2  and the N-channel transistor MN 2 .  
         [0112]    In both the push-pull transistors, because of the typical properties of the usual transistors, the forces at which the N-channel transistors MN 1 , MN 2  decrease (pull) the potentials of the output terminals C, CB are greater than the forces at which the P-channel transistors MP 1 , MP 2  increase (push) the potentials of the output terminals C, CB.  
         [0113]    Because of this fact, there may be the case that the (average) potentials of the output signals C, CB outputted from both the push-pull transistors are gradually decreased to thereby stop the oscillation of the VCO.  
         [0114]    So, in this embodiment, the current error compensation circuit  60  is mounted so as to keep the average values of the output terminals C, CB at the set values. As shown in FIG. 5, the average values of potentials of signals CQ 1 , CQ 2  and a referential voltage ref are inputted to a differential amplifier  61 . A signal based on a difference between those input signals is outputted to a current error compensation input terminal CMFBIN (refer to FIG. 4).  
         [0115]    Here, the signal CQ 1  has a voltage (capacitance terminal voltage) that results from a voltage drop through a resistor of the PLL filter  50  when the output signal C is inputted to the PLL filter  50 .  
         [0116]    Similarly, the signal CQ 2  has the voltage (capacitance terminal voltage) that results from the voltage drop through the resistor of the PLL filter  50  when the output signal CB is inputted to the PLL filter  50 .  
         [0117]    The detailed circuit configuration of the current error compensation circuit  60  will be described below with reference to FIG. 6.  
         [0118]    A signal outputted from an output terminal CMFBOUT of the current error compensation circuit  60  is inputted to the current error compensation input terminal CMFBIN in FIGS. 4, 5.  
         [0119]    The differential amplifier  61  is constituted by P-channel transistors MP 23 ,  24 ,  27  and  28 . The signal CQ 1  is inputted to a gate of the P-channel transistor MP 23 . The signal CQ 2  is inputted to a gate of the P-channel transistor MP 28 . The referential voltage ref is applied to the respective gates of the P-channel transistors MP 24 ,  27 .  
         [0120]    A drain of the P-channel transistor MP 21  for the constant current source is connected to respective sources of the P-channel transistors MP 23 ,  24 . A source of the P-channel transistor MP 21  for the constant current source is connected to the high potential side power supply AVDD.  
         [0121]    A drain of the P-channel transistor MP 25  for the constant current source is connected to respective sources of the P-channel transistors MP 27 ,  28 . A source of the P-channel transistor MP 25  for the constant current source is connected to the high potential side power supply AVDD.  
         [0122]    In a node NR, the referential voltage ref is set since the voltage between the high potential side power supply AVDD and the low potential side power supply AVSS is divided by resistors R 21 , R 22 , respectively.  
         [0123]    The operation of the current error compensation circuit  60  will be described below.  
         [0124]    In the differential amplifier  61 , each of the potentials of the signals CQ 1 , CQ 2  is compared with the referential voltage ref, and a signal on the basis of the difference is outputted to the output terminal CMFBOUT.  
         [0125]    A signal from the output terminal CMFBOUT is inputted to the current error compensation input terminal CMFBIN. Thus, the respective gate voltages of the N-channel transistor MN 11  for the constant current source and the N-channel transistor MN 17  for the constant current source are controlled. Hence, the value of the current is increased or decreased which flows through each of the N-channel transistor MN 11  for the constant current source and the N-channel transistor MN 17  for the constant current source.  
         [0126]    Here, each of the N-channel transistor MN 11  for the constant current source and the N-channel transistor MN 17  for the constant current source and the N-channel transistor MN 26  of the current error compensation circuit  60  constitute the current mirror circuit, as mentioned above.  
         [0127]    The N-channel transistor MN 11  for the constant current source and the N-channel transistor MN 17  for the constant current source are connected parallel to the N-channel transistor MN 10  for the constant current source and the N-channel transistor MN 16  for the constant current source, respectively. Thus, the values of the currents flowing through the differential transistors pair (MN 12 , MN 13 , MN 18 , MN 19 ) are controlled by the signal inputted to the current error compensation input terminal CMFBIN.  
         [0128]    As mentioned above, the current error compensation circuit  60  carries out the same phase signal feedback control (CMFB) so that the potentials of the signals CQ 1 , CQ 2  are equal to the referential voltage ref. Thus, there is no fear of the stop of the oscillation of the VCO.  
         [0129]    By the way, the current error compensation circuit  60  is effective not only for the property compensation of the above-mentioned push-pull transistor but also for the temperature compensation.  
         [0130]    According to this embodiment, the following effects can be obtained from the above-mentioned explanations.  
         [0131]    The large reduction in the switching noise of the charge pump suppresses the jitter caused by the switching noise.  
         [0132]    Since the current flowing into the charge pump is equal to the differential current, the residual current occurring in the conventional charge pump is never induced, which enables the suppression of the jitter caused by the residual current.  
         [0133]    The amount at which the peak current caused by a parasitic capacitance induced when an MOS transistor is switched is sent to the charge pump is reduced, which results in the suppression of the switching noise.  
         [0134]    It is possible to easily carry out the error current compensation of the charge pump that becomes trouble in the case of the differential control of the VCO. Thus, this enables the differential control type PLL to be stably attained using only the capacitance within a chip.  
         [0135]    According to the present invention, the switching noise is suppressed.