Abstract:
Display driver circuitry for electro-optic displays, in particular active matrix displays using organic light emitting diodes. The circuitry includes a driver to drive an electro-optic display element in accordance with a drive voltage, a photosensitive device optically coupled to the electro-optic display element to pass a current dependent upon illumination reaching photosensitive device, a control circuit having a control line coupled to the driver to control the brightness of the electro-optic display element and having a current sense input coupled to the photosensitive device, a current set line for coupling to a reference current generator, and a display element select line to, when active, cause the control circuit to drive the electro-optic display element in accordance with the current set by the reference current generator. The circuit provides improved control of an electro-display element such as an organic LED pixel.

Description:
This is the U.S. national phase of International Application No. PCT/GB02/04773 filed Oct. 23, 2002, the entire disclosure of which is incorporated herein by reference. 
   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   This invention generally relates to display drivers for electro-optic displays, and in particular relates to circuitry for driving active matrix organic light emitting diode displays. 
   2. Description of Related Technology 
   Organic light emitting diodes (OLEDs) comprise a-particularly advantageous form of electro-optic display. They are bright, colourful, fast-switching, provide a wide viewing angle and are easy and cheap to fabricate on a variety of substrates. Organic LEDs may be fabricated using either polymers or small molecules in a range of colours (or in multi-coloured displays), depending upon the materials used. Examples of polymer-based organic LEDs are described in WO 90/13148, WO 95/06400 and WO 99/48160; examples of so called small molecule based devices are described in U.S. Pat. No. 4,539,507. 
   A basic structure  100  of a typical organic LED is shown in  FIG. 1   a . A glass or plastic substrate  102  supports a transparent anode layer  104  comprising, for example, indium tin oxide (ITO) on which is deposited a hole transport layer  106 , an electroluminescent layer  108 , and a cathode  110 . The electro luminescence layer  108  may comprise, for example, a PPV (poly(p-phenylenevinylene)) and the hole transport layer  106 , which helps match the hole energy levels of the anode layer  104  and electroluminescent layer  108 , may comprise, for example, PEDOT:PSS (polystyrene-sulphonate-doped polyethylene-dioxythiophene). Cathode layer  110  typically comprises a low work function metal such as calcium and may include an additional layer immediately adjacent electroluminescent layer  108 , such as a layer of aluminium, for improved electron energy level matching. Contact wires  114  and  116  to the anode the cathode respectively provide a connection to a power source  118 . The same basic structure may also be employed for small molecule devices. 
   In the example shown in  FIG. 1   a  light  120  is emitted through transparent anode  104  and substrate  102  and such devices are referred to as “bottom emitters”. Devices which emit through the cathode may also be constructed, for example by keeping the thickness of cathode layer  110  less than around 50–100 nm so that the cathode is substantially transparent. 
   Organic LEDs may be deposited on a substrate in a matrix of pixels to form a single or multi-colour pixellated display. A multicoloured display may be constructed using groups of red, green, and blue emitting pixels. In such displays the individual elements are generally addressed by activating row (or column) lines to select the pixels, and rows (or columns) of pixels are written to, to create a display. It will be appreciated that with such an arrangement it is desirable to have a memory element associated with each pixel so that the data written to a pixel is retained whilst other pixels are addressed. Generally this is achieved by a storage capacitor which stores a voltage set on a gate of a driver transistor. Such devices are referred to as active matrix displays and examples of polymer and small-molecule active matrix display drivers can be found in WO 99/42983 and EP 0,717,446A respectively. 
     FIG. 1   b  shows such a typical OLED driver circuit  150 . A circuit  150  is provided for each pixel of the display and ground  152 , V ss    154 , row select  164  and column data  166  busbars are provided interconnecting the pixels. Thus each pixel has a power and ground connection and each row of pixels has a common row select line  164  and each column of pixels has a common data line  166 . 
   Each pixel has an organic LED  156  connected in series with a driver transistor  158  between ground and power lines  152  and  154 . A gate connection  159  of driver transistor  158  is coupled to a storage capacitor  160  and a control transistor  162  couples gate  159  to column data line  166  under control of row select line  164 . Transistor  162  is a field effect transistor (FET) switch which connects column data line  166  to gate  159  and capacitor  160  when row select line  164  is activated. Thus when switch  162  is on a voltage on column data line  166  can be stored on a capacitor  160 . This voltage is retained on the capacitor for at least the frame refresh period because of the relatively high impedances of the gate connection to driver transistor  158  and of switch transistor  162  in its “off” state. 
   Driver transistor  158  is typically an FET transistor and passes a (drain-source) current which is dependent upon the transistor&#39;s gate voltage less a threshold voltage. Thus the voltage at gate node  159  controls the current through OLED  156  and hence the brightness of the OLED. 
   The standard voltage-controlled circuit of  FIG. 1   b  suffers from a number of drawbacks. The main problems arise because the brightness of OLED  156  is dependent upon the characteristics of the OLED and of the transistor  158  which is driving it. In general, these vary across the area of a display and with time, temperature, and age. This makes it difficult to predict in practice how bright a pixel will appear when driven by a given voltage on column data line  166 . In a colour display the accuracy of colour representations may also be affected. 
   Two circuits which partially address these problems are shown in  FIGS. 2   a  and  2   b .  FIG. 2   a  shows a current-controlled pixel driver circuit  200  in which the current through an OLED  216  is set by setting a drain source current for OLED driver transistor  212  using a reference current sink  224  and memorising the driver transistor gate voltage required for this drain-source current. Thus the brightness of OLED  216  is determined by the current, I col′ , flowing into adjustable reference current sink  224 , which is set as desired for the pixel being addressed. It will be appreciated that one current sink  224  is provided for each column data line  210  rather than for each pixel. 
   In more detail, power  202 ,  204 , column data  210 , and row select  206  lines are provided as described with reference to the voltage-controlled pixel driver of  FIG. 1   b . In addition an inverted row select line  208  is also provided, the inverted row select line being high when row select line  206  is low and vice versa. A driver transistor  212  has a storage capacitor  218  coupled to its gate connection to store a gate voltage for driving the transistor to pass a desired drain-source current. Drive transistor  212  and OLED  216  are connected in series between a power  202  and ground  204  lines and, in addition, a further switching transistor  214  is connected between drive transistor  212  and OLED  216 , transistor  214  having a gate connection coupled to inverted row select line  208 . Two further switching transistors  220 , 222  are controlled by non-inverted row select line  206 . 
   In the embodiment of the current-controlled pixel driver circuit  200  illustrated in  FIG. 2   a  all the transistors are PMOS, which is preferable because of their greater stability and better resistance to hot electron effects. However NMOS transistors could also be used. This is also true of circuits according to the invention which are described below. 
   In the circuit of  FIG. 2   a  the source connections of the transistors are towards GND and for present generation OLED devices V ss  is typically around −6 volts. When the row is active the row select line  206  is thus driven at −20 volts and inverted row select line  208  is driven at 0 volts. 
   When row select is active transistors  220  and  222  are turned on and transistor  214  is turned off. Once the circuit has reached a steady state reference current I col′  into current sink  224  flows through transistor  222  and transistor  212  (the gate of  212  presenting a high impedance). Thus the drain-source current of transistor  212  is substantially equal to the reference current set by current sink  224  and the gate voltage required for this drain-source current is stored on capacitor  218 . Then, when row select becomes inactive, transistors  220  and  222  are turned off and transistor  214  is turned on so that this same current now flows through transistor  212 , transistor  214 , and OLED  216 . Thus the current through OLED is controlled to be substantially the same as that set by reference current sink  224 . 
   Before this steady state is reached the voltage on capacitor  218  will generally be different from the required voltage and thus transistor  212  will not pass a drain source current equal to the current, I col′ , set by reference sink  224 . When such a mismatch exists a current equal to the difference between the reference current and the drain-source current of transistor  212  flows onto or off capacitor  218  through transistor  220  to thereby change the gate voltage of transistor  212 . The gate voltage changes until the drain-source current of transistor  212  equals the reference current set by sink  224 , when the mismatch is eliminated and no current flows through transistor  220 . 
   The circuit of  FIG. 2   a  solves some of the problems associated with the voltage-controlled circuit of  FIG. 1   b  as the current through OLED  216  can be set irrespective of variations in the characteristics of pixel driver transistor  212 . However the circuit of  FIG. 2   a  is still prone to variations in the characteristic of OLED  216  between pixels, between active matrix display devices, and over time. A particular problem with OLEDs is a tendency for their light output to decrease over time, dependent upon the current with which they are driven (this may be related to the passage of electrons through the OLED). Such degradation is particularly apparent in a pixellated display where the relative brightness of nearby pixels can easily be compared. A further problem with the circuit of  FIG. 2   a  arises because each of transistors  212 ,  214  and  222  must be sufficiently physically large to handle the current through OLED  216 , which is equal to the I col  reference current. Large transistors are generally undesirable and, depending upon the active matrix device structure, may also obscure or prevent the use of part of a pixel&#39;s area 
   In an attempt to address these additional problems there have been a number of attempts to employ optical feedback to control the OLED current. These attempts are described in WO 01/20591, EP 0,923,067A, EP 1,096,466A, and JP 5-035,207 and all employ basically the same technique.  FIG. 2   b , which is taken from WO 01/20591, illustrates the technique, which is to connect a photodiode across the storage capacitor. 
     FIG. 2   b  shows a voltage-controlled pixel driver circuit  250  with optical feedback  252 . The main components of the driver circuit  250  of  FIG. 2   b  correspond to those of circuit  150  of  FIG. 1   b , that is, an OLED  254  in series with a driver transistor  256  having a storage capacitor  258  coupled to its gate connection. A switch transistor  260  is controlled by a row conductor  262  and, when switched on, allows a voltage on capacitor  258  to be set by applying a voltage signal to column conductor  264 . Additionally, however, a photodiode  266  is connected across storage capacitor  258  so that it is reverse biased. Thus photo diode  266  is essentially non conducting in the dark and exhibits a small reverse conductance depending upon the degree of illumination. The physical structure of the pixel is arranged so that OLED  254  illuminates photodiode  266 , thus providing an optical feedback path  252 . 
   The photocurrent through photodiode  266  is approximately linearly proportional to the instantaneous light output level from OLED  254 . Thus the charge stored on capacitor  258 , and hence the voltage across the capacitor and the brightness of OLED  254 , decays approximately exponentially over time. The integrated light output from OLED  254 , that is the total number of photons emitted and hence the perceived brightness of the OLED pixel, is thus approximately determined by the initial voltage stored on capacitor  258 . 
   The circuit of  FIG. 2   b  solves the aforementioned problems associated with the linearity and variability of the driver transistor  256  and OLED  254  but exhibits some significant drawbacks in its practical implementation. The main drawback is that every pixel of the display needs refreshing every frame as storage capacitor  258  is discharged over no more than this period. Related to this, the circuit of  FIG. 2   b  has a limited ability to compensate for ageing effects, again because the light pulse emitted from OLED  254  cannot extend beyond the frame period. Similarly, because the OLED is pulsed on and off it must be operated at an increased voltage for a given light output, which tends to reduce the circuit efficiency. Finally, capacitor  258  often exhibits non-linearities so that the stored charge is not necessarily linearly proportional to the voltage applied on column conductor  264 . This results in non-linearities in the voltage-brightness relationship for the pixel as photodiode  266  passes a photocurrent (and hence charge) which is dependent upon the level of illumination it receives. 
   SUMMARY 
   There is therefore a need for improved display driver circuitry for organic LEDs which addresses the above problems. 
   According to a first aspect of the present invention there is therefore provided display element driver circuitry for driving an element of an electro-optic display, the circuitry comprising, a driver to drive the electro-optic display element in accordance with a drive voltage, a photosensitive device optically coupled to the electro-optic display element to pass a current dependent upon illumination reaching the photosensitive device; and a control circuit having a control line coupled to the driver to control the brightness of the electro-optic display element and having a current sense input coupled to the photosensitive device, a current set line for coupling to a reference current generator and a display element select line to, when active, cause the control circuit to drive the electro-optic display element in accordance with a current set by the reference current generator. 
   Utilising optical feedback in this way allows the electro-optic display element light output to be directly controlled by a reference current flowing into a column line, and thus overcomes the problems associated with the prior art optical feedback technique in which the display element light output is effectively pulsed. Furthermore the linearity of the circuit&#39;s response is essentially controlled by the linearity of the photosensitive device and devices, which have good linearity, such as photo diodes, are relatively easy to fabricate. As will be explained below, the circuit also needs only one large transistor, for the driver, rather than the three large transistors required by a current-controlled driver circuit in which the drive current rather than the light output is servoed. 
   Preferably the display driver circuitry includes a storage element, such as a capacitor or digital capacitor, coupled to the control line. In this way, when the element select line is inactive a drive voltage set by the reference current generator may be memorised. 
   The storage element may comprise an internal capacitance of the driver and, where the driver comprises a FET (Field Effect Transistor) the storage element may simply comprise the FET gate capacitance. The FET may be fabricated for increased gate capacitance to effectively integrate the storage element with the driver transistor. In use an error current flows into or out of the control line to deposit or remove charge from the capacitor, to change the voltage across capacitor and hence the drive voltage. 
   In a preferred embodiment a common-gate (FET) transistor or common-base (bipolar) transistor is coupled between the photosensitive device and the current sense input to reduce the voltage across the photosensitive device. Reducing the voltage across the device reduces the leakage current through the device, which is advantageous because the photocurrent through the device is generally relatively small, particularly at low display element brightness levels. This common-gate or common-base transistor may advantageously be biased using a second transistor with a matched V T  (gate-source threshold voltage) or a matched V be  (base-emitter voltage). Current can then be passed through the second transistor to set a gate (or base) voltage for the second transistor which can then be applied to the common-gate (or common-base) transistor to set an appropriate bias point. 
   In a refinement of this preferred embodiment the reference current flowing in the column line may be diverted through the second transistor in an initial bias-set cycle before the optical feedback path is utilized. This may be achieved by providing a switch to divert the current through the second transistor and, preferably, a second switch and a further storage element to hold a bias condition set in this way. The switches are preferably controlled by a compensate line which is activated to set the bias for the common-gate (or common-base) transistor before the display element select line is activated. 
   In a preferred embodiment display element driver circuitry of the above-described type is provided for each pixel in an active matrix display. In such an arrangement a display, row address line is coupled to the display element select lines of pixels in a corresponding row, and a display element column select line is coupled to the current set lines of pixels in a corresponding column, or vice-versa. A programmable reference current generator is then preferably provided for each column address line so that the brightness of pixels in a selected row may be programmed. 
   In a corresponding aspect the invention also provides a method of controlling the brightness of electro-optic display elements in an active matrix display, the method comprising, providing a photosensitive device for each element, the photosensitive device passing a photocurrent dependent upon the illumination of the device, sensing the brightness of each element by sensing the photocurrent passed by the photosensitive device for the element; and controlling the brightness of each element so that the sensed photocurrent is determined by and preferably substantially matches a reference current. 
   Preferably the active matrix display includes a voltage-controlled driver for each display element, each driver having a storage capacitor to store a display element drive voltage. The method may then further comprise compensating for a difference between the reference current and the photocurrent by charging or discharging the storage capacitor. 
   As described above the method preferably further includes operating the photosensitive device under reduced bias conditions by dropping at least a portion of a bias voltage for the device across a transistor. In refinement of this method a bias cycle is provided prior to the brightness sensing and controlling, to set a bias for the photosensitive device using the reference current. 
   Preferably the electro-optic display element comprises an organic light emitting diode. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and other aspects of the invention will now be further described by way of example only, with reference to the accompanying figures in which: 
       FIGS. 1   a  and  1   b  show, respectively, a basic organic LED structure, and a typical voltage-controlled OLED driver circuit; 
       FIGS. 2   a  and  2   b  show, respectively, a current-controlled OLED driver circuit, and a voltage-controlled OLED driver circuit with optical feedback according to the prior art; 
       FIGS. 3   a  to  3   c  show, respectively, a current-controlled OLED driver circuit with optical feedback, a first alternative switching arrangement, and a second alternative switching arrangement; 
       FIG. 4  shows a current-controlled OLED driver circuit with optical feedback and reduced photodiode bias; 
       FIG. 5  shows a current-controlled OLED driver circuit with optical feedback and photodiode bias nulling means; and 
       FIGS. 6   a  and  6   b  show vertical cross sections through device structures of OLED display elements with driver circuits incorporating optical feedback. 
   

   DETAILED DESCRIPTION 
   Referring first to  FIG. 3   a , this shows a current-controlled organic LED driver circuit  300  with optical feedback according to an embodiment of the present invention. In an active matrix display typically each pixel is provided with such a driver circuit and further circuitry (not shown) is provided to address the pixels row-by-row, to set each row at the desired brightness. To power and control the driver circuitry and OLED display element such an active matrix display is provided with a grid of electrodes including, as shown, a ground (GND) line  302 , a power or V ss  line  304 , a row select line  306  and a column data line  308 . Each column data line is connected to a programmable constant current reference source (or sink)  324 . This is not part of the driver circuitry provided for each pixel but instead comprises part of the circuitry provided for each column. Reference current generator  324  is programmable so that it can be adjusted to a desired level to set a pixel brightness, as described in more detail below. 
   The driver circuit  300  comprises a driver transistor  310  connected in series with an organic LED display element  312  between the GND  302  and V ss    304  lines. A storage capacitor  314 , which may be integrated with the gate of transistor  310 , stores a charge corresponding to a memorised gate voltage to control the drive current through OLED element  312 . Control circuitry for the driver comprises two switching transistors  320 ,  322  with a common gate connection coupled to row select line  306 . When row select line  306  is active these two switch transistors are on, that is the switches are “closed”, and there is a relatively low impedance connection between lines  315 ,  317  and  308 . When row select line  306  is inactive transistors  320  and  322  are switched off, capacitor  314  and the gate of transistor  310  are effectively isolated, and any voltage set on capacitor  314  is memorised. 
   In the circuit of  FIG. 3   a , and in the circuits of  FIG. 3   b ,  3   c ,  4  and  5  described later, the transistors are all PMOS. 
   A photodiode  316  is coupled between GND line  302  and line  317  so that it is reverse biassed. The photodiode is physically arranged with respect to the OLED display element  312  such that an optical feedback path  318  exists between OLED  312  and photodiode  316 . In other words, OLED  312  illuminates photodiode  316  and this allows an illumination-dependent current to flow in a reverse direction through photodiode  316 , that is from GND line  302  towards V ss . As the skilled person will understand, broadly speaking each photon generates an electron within photodiode  316  which can contribute to a photocurrent. 
   Column data line  308  is coupled, at the end of a column, to programmable reference current generator  324 . This attempts to cause a reference current, which will be referred to as I col , to flow to off-pixel V ss  connection  326 . Line  317  may be referred to as a current sense line, passing a current I sense  and line  315  may be referred to as a control line, passing a current I error  to set a voltage on capacitor  314  to control OLED  312 . When row select line  306  is active and transistors  320  and  322  are I col =I sense +I error  and thus a current I error  flows either onto or off capacitor  314  until OLED  312  illuminates photodiode  316  such that I sense =I col . At this point row select line  306  can be deactivated, and the voltage required for this level of brightness is memorised by capacitor  314 . 
   The time required for the voltage on capacitor  314  to stabilise depends upon a number of factors, which may be varied in accordance with the desired device characteristics, and may be a few microseconds. Broadly speaking a typical OLED drive current is of the order of 1 μA whilst a typical photocurrent is around 0.1% of this, or of the order of 1 nA (in part dependent upon the photodiode area). It can therefore be seen that the power handling requirements of transistors  320  and  322  are negligible compared with that of the drive transistor  310 , which must be relatively large. To speed up the settling time of the circuit it is preferable to use a relatively small value for capacitor  314  and a relatively large area photodiode to increase the photocurrent. This also helps reduce the risk of noise and stability at very low brightness levels associated with stray or parasitic capacitance on column data line  308 . 
     FIGS. 3   b  and  3   c  show a portion of the circuit of  FIG. 3   a  illustrating different possible configurations for switching transistors corresponding to switching transistors  320  and  322  of  FIG. 3   a  The purpose of transistors  320  and  322  is to couple lines  315 ,  317  and  308  when row select line  306  is active and it will be appreciated that there are three different ways of connecting three nodes using two controllable switches. In  FIG. 3   b  a first switching transistor  350  is connected between lines  308  and  315  and a second switching transistor  352  is connected between lines  315  and  317 . Both transistors  350  and  352  are controlled by row select line  306 . In  FIG. 3   c  a first switching transistor  360  is connected between lines  308  and  315  and a second switching transistor  362  is connected between lines  308  and  317 . Optionally a third switching transistor  364  may be connected between lines  315  and  317 . The two (or three) switching transistors are all controlled by row select line  306 . 
   One drawback of the basic circuit of  FIG. 3   a  is the leakage current through photodiode  316  which flows when this photodiode is reverse biased. The leakage current is voltage dependent and thus it can be reduced by reducing the bias voltage across photodiode  316 .  FIG. 4  shows an improved circuit  400  in which this is achieved. The circuit of  FIG. 4  is a modification of the circuit of  FIG. 3   a  and elements indicated by reference numerals  402  to  426  correspond to elements  302  to  326  of the circuit of  FIG. 3   a.    
   The additional components in driver circuit  400  of  FIG. 4 , as compared with driver circuit  300  of  FIG. 3   a , are transistors  428  and  430  and resistor  432 . In driver circuit  300  of  FIG. 3   a  when row select  306  is active the voltage across photodiode  316  is approximately equal to the gate voltage of driver transistor  310  on line  315 , because switching transistor  320  is on (closed). As the skilled person will be aware, the gate voltage on a FET is equal to a threshold voltage V T , plus an additional voltage, which will be referred to as V control , required to set the desired drain-source current, I ds . In  FIG. 4  transistor  428  is used to drop at least this threshold voltage, thus leaving only a voltage approximately equal to V control  across photodiode  416 . This is done by employing transistor  428  in a common-gate configuration, with a gate bias voltage set by transistor  430  and resistor  432 . 
   In the embodiment drawn in  FIG. 4  transistors  428  and  430  are both PMOS devices and so have their source connection towards GND. Transistor  430  has its drain and gate coupled together and thus operates as a (non-linear) resistor. Transistor  430  is connected in series with resistor  432  between GND line  402  and V ss  line  404 , and a drain-source current of transistor  430  is determined by the transistor characteristics and the value of resistor  432 . The gate voltage of transistor  430  necessary to provide this drain-source current is equal to the gate threshold voltage for transistor  430  plus an additional control voltage. The gate of transistor  428  is coupled to the gate of transistor  430  so that their gate voltages are substantially the same. Transistors  428  and  430  are preferably both matched so that they have substantially the same threshold voltage. 
   From the foregoing explanation it will be appreciated that transistor  428  drops an FET threshold voltage plus a small additional control voltage dependent upon the drain-source current of transistor  430  set by resistor  432 . When transistor  420  is on the voltage on line  417  is approximately equal to that on the gate of transistor  410 . The threshold voltages of transistors  410  and  428  are approximately the same so that the bias voltage on photodiode  416  will therefore be approximately equal to the difference in V control  on the gate of transistor  410  and on the gate of transistor  430 . Preferably the drain-source current of transistor  430  is chosen to be similar to the drain-source current of transistor  410  when OLED  412  is dimly illuminated. 
   In operation the photocurrent I sense  in line  417  is substantially unchanged as there is no alternative path for the current to take. Thus the servo mechanism of transistors  420  and  422  operates in the same way as the servo mechanism of transistors  320  and  322  in driver circuit  300 . Transistor  428  is largely off, being turned on by an amount dependent upon the photocurrent through photodiode  416 . As with driver circuit  300  capacitor  414  is charged such that this photocurrent, I sense , equals I col . 
   Some exemplary but not necessarily typical voltage values can be used to illustrate how the circuit works in practice. When OLED  412  is dark a voltage across photodiode  416 , V pd  equals −1 volt say, transistor  428  is substantially off, and the gate source voltage of transistor  428 , V GS  is ≅V T . When OLED  412  is dimly lit, V PD  equals −0.9 volt say, transistor  428  is slightly on and V GS ≅V T +0.1V. When OLED  412  is bright V PD  equals −0.5 volt say, transistor  428  is on, and V GS ≅V T +0.5v. When photodiode  416  is extremely brightly illuminated the photodiode may operate as a photocell, in which case V PD  equals +0.2 volt say, transistor  428  is full on, and V GS ≅V T +1.2v. 
   The circuit of  FIG. 4  helps to reduce inaccuracies caused by leakage current through the photodiode by dropping approximately V T  across transistor  428 , but still leaves a residual photodiode bias voltage roughly corresponding to the (variable) control voltage required in addition to V T . Thus the photo diode bias changes with the desired brightness of OLED  412 —the brighter the OLED the less the reverse bias—in effect due to the finite transconductance of transistor  428 . Employing a bipolar transistor rather than a FET for transistor  428  would increase the transconductance but reduce the accuracy with which I col  determines I sense .  FIG. 5  shows a circuit in which the reference current I col  can be directed through a bias set transistor to effectively null out this additional variation in photodiode bias voltage. 
   Referring to  FIG. 5 , this shows a driver circuit  500  including means to null a photodiode bias voltage. The driver circuit  500  of  FIG. 5  is a modification of the driver circuit  400  of  FIG. 4  and elements  502  to  530  correspond to elements  402  to  430  in  FIG. 4 . However resistor  432  coupling the drain of transistor  430  to V ss  has been replaced by a transistor  534  coupling the drain of transistor  530  to column data line  508  via connection  540 . The link between the drain and gate of transistor  430  has been broken and transistor  532  is now connected between the drain and gate of transistor  530 . A bias voltage hold capacitor  536  has also been connected to the coupled gates of transistors  528  and  530 . Transistors  532  and  534  operate as FET switches controlled by compensate line  538 . 
   When compensate line  538  is active transistors  532  and  534  are switched on. The driver circuit  500  then operates in a similar manner to driver circuit  400 , except that when row select line  506  is inactive the drain-source current of transistor  530  is substantially equal to the reference current, I col , flowing into current sink  524 , as transistor  522  is off. Thus when compensate line  538  is active and row select line  506  is inactive the gate voltage of transistor  530  is equal to the gate threshold voltage of transistor  530  plus the additional control voltage needed to provide a drain-source current in transistor  530  equal to I col . Preferably transistor  530  is substantially matched to transistor  528  so that when the drain source current of transistor  528  is equal to I col  and the gate source voltage of transistor  528  is the same as the gate source voltage of transistor  530  substantially all the photodiode bias voltage is dropped across transistor  528  leaving substantially zero bias voltage across photodiode  516 . Capacitor  536  is connected to the gates of transistors  528  and  530  to store the bias voltage set in this way. 
   The driver circuit  500  of  FIG. 5  is operated in two stages, a first, bias cycle stage in which a bias voltage is set for transistor  528  via transistor  530 , and a second, pixel control stage in which the brightness of OLED  512  is controlled according to the reference current I col . In the bias cycle stage compensate line  538  is active and row select line  506  is inactive; in the pixel control stage row select line  506  is active and compensate line  538  is inactive. Initially, compensate line  538  is activated and row select line  506  is deactivated for a predetermined interval, to allow capacitor  536  to be charged to the required bias voltage. Compensate line  538  is then deactivated and row select line  506  is activated and the main optical feedback servo loop is allowed to stabilise over a second predetermined interval. Both intervals are typically of the order of one to a few microseconds. Row select line  506  is then deactivated, capacitor  514  maintaining OLED  512  at its set brightness. 
   Referring now to  FIG. 6 , this shows, in outline, two alternative physical structures for OLED pixel driver circuits incorporating optical feedback (the drawings are not to scale).  FIG. 6   a  shows a bottom-emitting structure  600  and  FIG. 6   b  shows a top-emitter  650 . 
   In  FIG. 6   a  an OLED structure  606  is deposited side-by-side with polysilicon driver circuitry  604  on a glass substrate  602 . The driver circuitry  604  incorporates a photodiode  608  to one side of the OLED structure  606 . Light  610  is emitted through the bottom (anode) of the substrate. 
     FIG. 6   b  shows a cross section through an alternative structure  650  which emits light  660  from its top (cathode) surface. A glass substrate  652  supports a first layer  654  comprising the driver circuitry and including a photodiode  658 . An OLED pixel structure  656  is then deposited over the driver circuitry  654 . A passivation or stop layer may be included between layers  654  and  656 . Where the driver circuitry is fabricated using (crystalline) silicon rather than polysilicon or amorphous silicon a structure of the type shown in  FIG. 6   b  is required and substrate  652  is a silicon substrate. 
   In the structures of  FIGS. 6   a  and  6   b  the pixel driver circuitry may be fabricated by conventional means. The organic LEDs may be fabricated using either ink jet deposition techniques such as those described in EP 880303 to deposit polymer-based materials or evaporative deposition techniques to deposit small molecule materials. Thus, for example, so-called micro-displays with a structure of the type illustrated in  FIG. 6   b  may be fabricated by ink jet printing OLED materials onto a conventional silicon substrate on which CMOS pixel driver circuitry has previously been fabricated. 
   The illustrated embodiments of the driver circuit use PMOS transistors but the circuits may be inverted and NMOS may be employed or, alternatively, a combination of PMOS and NMOS transistors may be used. The transistors may comprise thin film transistors (TFTs) fabricated from amorphous or poly-silicon on a glass or plastic substrate or conventional CMOS circuitry may be used. In other embodiments plastic transistors such as those described in WO 99/54936 may be employed, and the photodiode may comprise a reverse biased OLED to allow the entire circuitry to be fabricated from plastic. Similarly although the circuit has been described with reference to field effect transistors, bipolar transistors may also be used. 
   The display element driver circuitry has been described with reference to its use for driving organic LEDs but the circuitry may also be employed with other types of electroluminescent display such as inorganic TFEL (Thin Film Electroluminescent) displays, gallium arsenide on silicon displays, porous silicon displays, photoluminescence quenching displays as described in UK patent application no. 0121077.2, and the like. Although the driver circuitry primarily finds applications in active matrix displays it may also be used with other types of display such as segmented displays and hybrid semi-active displays. 
   The preferred photosensor is a photodiode which may comprise a PN diode in TFT technology or a PIN diode in crystalline silicon. However other photosensitive devices such as photoresistors and photosensitive bipolar transistors and FETs may also be employed, providing they have a characteristic in which a photocurrent is dependent upon their level of illumination. 
   No doubt many other effective alternatives will occur to the skilled person and it should be understood that the invention is not limited to the described embodiments.