Abstract:
Described is a compact, lower power gated ring oscillator time-to-digital converter that achieves first order noise shaping of quantization noise using a digital implementation. The gated ring oscillator time-to-digital converter includes a plurality of delay stages configured to enable propagation of a transitioning signal through the delay stages during an enabled state and configured to inhibit propagation of the transitioning signal through the delay stages during a disabled state. Delay stages are interconnected to allow sustained transitions to propagate through the delay stages during the enabled state and to preserve a state of the gated ring oscillator time-to-digital converter during the disabled state. The state represents a time resolution that is finer than the delay of at least one of the delay stages. A measurement module determines the number of transitions of the delay stages.

Description:
RELATED APPLICATIONS 
       [0001]    This application claims the benefit of the earlier filing dates of U.S. Provisional Patent Application Ser. No. 60/844,742, filed Sep. 15, 2006, titled “Method and Apparatus for Efficient Scrambling and/or Frequency Noise Shaping of Time-to-Digital Converter Quantization Noise”, the entirety of which is incorporated herein by reference. 
     
     FIELD OF THE INVENTION 
       [0002]    The present invention relates generally to time-to-digital converters. More particularly, the invention relates to a gated ring oscillator for frequency noise shaping of time-to-digital quantization noise. 
       BACKGROUND OF THE INVENTION 
       [0003]    New generations of CMOS IC processes provide faster transistors with greater circuit density; however, many traditional analog characteristics are sacrificed. Consequently, circuit designers have pursued new mixed-signal circuit architectures that take advantage of digital circuits to augment analog processing of signals. In particular, digital phase locked loops (PLLs) have benefited from the use of time, rather than voltage or current, as a signal domain. 
         [0004]    A time-to-digital converter (TDC) is used to quantify time information of a signal event with respect to a reference event. Traditionally, TDCs have been used in applications in experimental physics and laser range finding. More recently, fully integrated TDCs have been investigated as a core building block for a variety of clocking and phase-locked loop systems and applications. 
         [0005]    A typical TDC operates as shown in  FIG. 1 . The time difference t in  to be measured is defined by the difference between a reference signal edge occurring at a time t ref  and an event signal edge occurring at a time t event . To obtain a quantitative measurement of t in , time is segmented into discrete steps according to the timing resolution t res  of the TDC. A measured time difference t out  is provided by the TDC to approximate t in  with some quantization error t q , where t q =t out −t in . 
         [0006]    A TDC  10  is commonly implemented with standard digital circuit elements as shown in  FIG. 2 . The reference signal is propagated through a serial configuration of delay stages  14 A to  14 J (generally  14 , only three shown for clarity). In time, the reference signal edge moves from left to right in the figure, from the first delay stage  14 A to the last delay stage  14 J. At a rising event signal edge, the states of all the delay elements  14  are clocked into respective serial registers  18 A to  18  K (generally  18 ). The output time t out  of the TDC  10  is determined according to the product of a positive integer output N out  and the delay t res  of a single delay stage  14 . Thus the minimum resolution of the TDC  10  is limited by process technology. The quantization error of the TDC has a substantially white spectrum although practical applications can exhibit quantization noise with significant tone content for certain input signals. 
         [0007]    A vernier delay TDC is based on a variation of the conventional TDC structure and achieves an improvement in the resolution of the TDC. An example of a vernier delay TDC  20  is shown in  FIG. 3 . The reference signal edge propagates through the serial configuration of delay elements  14 . The event signal edge propagates through a separate serial configuration of delay elements  22 A to  22 J (generally  22 ). The delay t delay  caused by each delay element  14  in the propagation path for the reference signal edge is slightly greater by a time t offset  than the delay of each delay element  22  in the propagation path for the event signal edge. Thus, if the reference signal edge occurs first and the two edges propagate through their respective delay elements  14  and  22 , the event signal edge slowly catches up to the reference signal edge. Ideally, the vernier delay TDC  20  improves the time resolution by t delay /t offset . Mismatches within the propagation paths and thermal noise contributed by the registers  18  limits the minimum TDC resolution. In addition, the number of delay elements  14  and  22  required for a maximum acceptable input time difference increases compared to a standard TDC, thereby reducing throughput and making the vernier delay TDC  20  unsuitable for applications requiring a large dynamic range. 
         [0008]    The noise performance of a TDC can be an obstacle for meeting the noise requirements of many applications, including applications based on current wireless communications standards. To reduce the effect of a high TDC resolution, the bandwidth of a PLL can be reduced. The reduced bandwidth sacrifices utility such as the option to modulate a data stream within the PLL that is not otherwise possible without a significant reduction in the TDC resolution. Improved TDC resolution is not likely to be realized soon through processing technology advances and the vernier delay TDC  20  is not a practical option for many applications due to its inherent circuit complexity. 
         [0009]    What is needed is a TDC that overcomes the problems and limitations described above. The present invention satisfies this need and provides additional advantages. 
       SUMMARY OF THE INVENTION 
       [0010]    In one aspect, the invention features a gated ring oscillator time-to-digital converter for modifying time-to-digital converter quantization noise. The gated ring oscillator time-to-digital converter includes a plurality of delay stages configured to enable propagation of a transitioning signal through the delay stages during an enabled state and configured to inhibit propagation of the transitioning signal through the delay stages during a disabled state. The delay stages are interconnected to allow sustained transitions to propagate through the delay stages during the enabled state and to preserve a state of the gated ring oscillator time-to-digital converter during the disabled state. The state represents a time resolution that is finer than the delay of at least one of the delay stages. The gated ring oscillator time-to-digital converter also includes a measurement module to determine a number of transitions of the delay stages. 
         [0011]    In another aspect, the invention features a method for determining a measurement time. The measurement time has a plurality of enable periods to be included in the determination and at least one disable period to be excluded from the determination. The method includes enabling propagation of a transitioning signal through a plurality of delay stages at a first enable time. The delay stages are interconnected to allow sustained transitions to propagate through the delay stages during the enabled state and preservation of a state of the delay stages during a disabled state. The propagation of the transitioning signal through the delay stages is disabled at a disable time and the state of the delay stages at the disable time is stored. Propagation of the transitioning signal through the delay stages is enabled at a second enable time. The propagation resumes at the state of the delay stages at the disable time. The state represents a time resolution that is finer than the delay of at least one of the delay stages. 
     
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0012]    The above and further advantages of this invention may be better understood by referring to the following description in conjunction with the accompanying drawings, in which like numerals indicate like structural elements and features in the various figures. For clarity, not every element may be labeled in every figure. The drawings are not necessarily to scale, emphasis instead being placed upon illustrating the principles of the invention. 
           [0013]      FIG. 1  shows the timing relationship for a typical TDC. 
           [0014]      FIG. 2  is a diagram of a conventional TDC employing standard digital circuit elements. 
           [0015]      FIG. 3  is a diagram of a vernier delay as is known in the art. 
           [0016]      FIG. 4  illustrates an embodiment of a TDC according to the invention. 
           [0017]      FIG. 5  illustrates a digital inverter that can be used for each delay element in the gated ring oscillator of  FIG. 4 . 
           [0018]      FIG. 6  shows a timing diagram for operation of the TDC of  FIG. 4 . 
           [0019]      FIG. 7  illustrates another embodiment of a TDC in accordance with the invention. 
           [0020]      FIG. 8  illustrates a buffer and a regenerative latch that can be used between the output of each delay element and counter of the TDC of  FIG. 7 . 
           [0021]      FIG. 9A  shows how a signal edge propagates through a series of inverters in a gated ring oscillator and  FIG. 9B  shows how the time difference t out −t en  in  FIG. 9A  is a function of the disabled state of the gated ring oscillator. 
           [0022]      FIG. 10  shows a DC transfer curve for a gated ring oscillator. 
           [0023]      FIG. 11A  shows a quantized signal in the time domain and  FIG. 11B  shows the frequency spectrum of the signal of  FIG. 11A . 
           [0024]      FIG. 12A  shows a gated ring oscillator for an embodiment of a TDC according to the invention and  FIG. 12B  illustrates a portion of the gated ring oscillator of  FIG. 12A  implemented in a PMOS and NMOS transistor configuration. 
       
    
    
     DETAILED DESCRIPTION 
       [0025]    In brief overview, the invention relates to gated ring oscillator TDC for modifying time-to-digital converter quantization noise. The spectral density of the quantization noise of the TDC is substantially reduced relative to classical TDC architectures with the same timing resolution. The TDC can be fabricated as a small area of an integrated circuit having low power requirements. Products requiring high timing accuracy can realize significant advantages with the TDC of the present invention. By way of example, such products include wireless communication devices (e.g., cell phones and local area networks), field programmable gate arrays, microprocessors, microcontrollers and digital signal processors, high-speed application specific integrated circuits (ASICs), routers and high-speed serial links. 
         [0026]      FIG. 4  illustrates an embodiment of a TDC  30  according to the invention. The TDC  30  includes a gated ring oscillator (GRO)  34  and a measurement module  38 . The GRO  34  includes a plurality of delay elements  42 A,  42 B and  42 C (generally  42 , only three shown for clarity) illustrated as digital inverters although other forms of delay elements are contemplated. Unlike TDCs employing conventional oscillator structures which lose their state information for inactive gaps in the time measurement window, the GRO  34  of the illustrated TDC  30  counts the oscillation cycles during the measurement time and holds the state of the oscillator between measurements. By holding the residual output state during the inactive gaps of a longer measurement window, the next measurement “starts in the same place” that the previous measurement ended. The resulting quantization steps are effectively smaller than are possible using standard gate delays. Advantageously, due to the scrambling achieved by mismatch of the delay elements  42  and noise within the GRO  34 , the TDC  30  achieves first-order noise shaping and has reduced tone content compared to conventional TDC architectures. 
         [0027]    Ideally, upon enabling the GRO  34  for a measurement, oscillation begins immediately at a nominal frequency and, upon disabling the GRO  34 , the frequency instantaneously returns to zero and the state of each delay stage in the GRO  34  is preserved (i.e., held at a constant value). A practical GRO, however, has a finite transition time when going between nominal frequency and zero frequency, and a key design goal is to minimize that transition time in order to reduce the impact of noise. Another practical issue is that the GRO delay stages are susceptible to non-ideal circuit issues such as charge injection, leakage currents, and noise. Such effects can cause transitions to effectively occur after the GRO is disabled despite best efforts in preserving the state of each delay stage. To accommodate the presence of such “disable mode” transitions, the transition measurement module  38  is designed to include these “disable mode” transitions in the overall transition count so that additional measurement noise is avoided and noise shaping of the time-to-digital converter quantization noise is maintained. 
         [0028]      FIG. 5  shows an example of a digital inverter  46  that can be used for each delay element  42  in the GRO  34 . The digital inverter  46  includes two NMOS transistors  50  (M 1  and M 2 ) and two PMOS transistors  54  (M 3  and M 4 ). The propagation path A is coupled to the gate of the M 2  NMOS transistor  50  and the gate of the M 3  PMOS transistor  54 . The digital inverter  46  is gated according to an enable signal and a complementary enable signal applied to the gates of the M 1  NMOS transistor  50  and the M 4  PMOS transistor  54 , respectively. 
         [0029]    Referring also to  FIG. 4 , the GRO  34  ideally turns on and off in response to the enable signal ENABLE which has a pulse width equal to the particular time interval to be measured. The quantized output is determined from the number of edge transitions of the individual digital inverters  42  for each enabled time interval. For reduced clocking speeds, a larger number of delay elements are used to permit ample time for resetting the clocked registers in the measurement module  38 . 
         [0030]    The timing diagram of  FIG. 6  shows one example of operation of the TDC  30 . The states TAP 0 , TAP 1  and TAP 2  of the digital inverters  42 , the measured time difference, and the quantization error are shown as a function of time relative to the edges to be measured. During an enabled state of operation, the output state of each digital inverter  42  is a function of its input state and transient switching characteristics. During the disabled state of operation, operation of the digital inverters  42  is suspended and the output state of each digital inverter  42  is ideally held at a constant value. Referring also to  FIG. 5 , if the leakage currents through the transistors  50  and  54  are negligible during the disabled state, the output states of the digital inverters  42  do not change because the current path is removed from the output and charge is conserved on the output capacitance (not shown). Thus the phase of the GRO  34  is preserved during the disabled time and phase integration resumes when the enable signal is reasserted. As a result, the residual phase (as shown as the shaded region at the end of each sample) is carried over to the next quantized phase sample (as shown by the shaded region at the beginning of the each subsequent sample). In effect, the common component of residual error between the current and previous sample is canceled to achieve first-order noise shaping of the quantization noise in the quantized time measurement. 
         [0031]    In an alternative embodiment, differential operation is used to minimize rise and fall time mismatches by limiting clocking to a positive edge transition or a negative edge transition. In addition, the delay elements are not limited to a highly-digital form such as digital inverters. Rather, the delay elements need only to provide a delay and to have the ability to be enabled and disabled in such a way that state information is preserved during the inactive measurement gaps. The delay can therefore be either fixed or variable according to the system requirements. 
         [0032]    Various modifications can be used to further improve performance of the TDC of the invention. For example, the time interval of interest can be altered, linearly or otherwise, before application of the timing signals or the enable signal to the GRO to obtain improved quantization noise performance. 
         [0033]      FIG. 7  is a schematic illustration of a GRO  60  used in another embodiment of a TDC in accordance with the invention. The GRO  60  includes a plurality of delay elements (shown as digital inverters  42 ) arranged serially in a ring configuration. The output of each delay element  42  is in communication with a measurement module that observes the state of the GRO  60  and provides a quantized time output. Although the measurement module can be configured in a number of ways, in this case each delay element  42  is in communication with a quantization latch  64  which in turn is in communication with a counter  68  having reset capability. Each counter  68  is coupled through a respective summing node or “adder”  72 . The adders  72  are serially connected to a quantized time output node  76 . 
         [0034]    When the GRO  60  is enabled, the outputs of the delay elements  42  are provided to the counters  68  through a transparent latch  80 . Each counter  68  determines a counter value that indicates the number of positive and negative edge transitions for its delay element  42 . When the GRO  60  is disabled, each counter value is registered into its adder  72  to be summed with the other counter values. The counters  68  are reset to zero before the next time the GRO  60  is enabled. One or more of the delay element  42  is in transition when the oscillation is suspended, therefore changes above and below the quantization threshold can be mistaken by the corresponding counters as a GRO transition. To prevent this occurrence, a buffer  78  and a regenerative transparent latch  80  are provided between the output of each delay element  42  and its counter  68  as shown in  FIG. 8 . The buffer  78  provides gain and reverse isolation. The latch  80  restricts the time window during which the signal applied to a counter  68  can cause a transition and also ensures a quantization decision through positive feedback. During the disabled periods, if there is change at the output of any of the delay elements  42  due to charge injection, drift, leakage, or noise, the counter values are not corrupted. Upon subsequent enablement of the GRO  60 , any delay element  42  in transition is allowed to complete its transition before the latch  80  is activated such that the counter  68  only observes a single transition event for a single transition of the delay element  42 . 
         [0035]    The GRO as described in the embodiments above can be limited by distortion of the residual error that occurs when the GRO is disabled and enabled. In particular, the time for the disabled GRO to resume oscillation upon an enable assertion depends in a non-linear manner on the voltages, i.e., the internal state, of the GRO stored during the disabled time. Due to this distortion, the GRO may be unable to accurately transfer the oscillation state from one sample to the next, thus adversely affecting the noise shaping. 
         [0036]    The key issue is that propagation delay of the edges through the different stages of the ring oscillator is impacted by the gating operation, which then corrupts the state of the oscillator such that its starting point in a new measurement cycle does not match the stopping point of the previous measurement cycle. The resulting state corruption, in turn, prevents proper progression of the ring oscillator edges through their respective delay stages as the GRO is repeatedly gated on and off, and thereby leads to dead zone behavior. 
         [0037]    To illustrate this behavior,  FIG. 9A  shows a GRO  84  where a signal edge (shown as “Input”) propagates through a series of inverters  42 . The time at which the GRO  84  is disabled is independently defined as t en , and the time at which the last inverter undergoes an edge transition (shown as “Output”) is t out . Ideally, the time difference t out −t en  is a constant value as graphically described by the correspondence between the Input edge and the Output edge, as this means that the GRO startup time is constant. As shown in  FIG. 9B , the time difference t out −t en  is not constant, but is instead a function of the disabled state of the GRO  84 . The residual error distortion is a complex function of non-linear transistor behavior across both rising and falling edges, and accounts for the period of the illustrated time difference function being twice the delay caused by a delay element  42 . 
         [0038]    One effect of the distortion are the “deadzones” in the DC transfer curve shown in  FIG. 10  where the normalized digital output 
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         [0000]    of a TDC is shown as a function of a normalized input signal 
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         [0000]    For input signal levels approximately equal to a single delay element, a GRO is unable to distinguish small changes in time with respect to the Input edge (see  FIG. 9 ) so that the TDC simply provides an integer value without any noise shaping behavior. Deadzones in the illustrated transfer curve are evident near and at integer values of the normalized input signal 
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         [0000]    The effect of this distortion can also be observed in the frequency domain.  FIG. 11A  shows a quantized signal having a fundamental frequency of approximately 75 KHz in the time domain. The non-linearity is not readily observable; however,  FIG. 11B  shows the frequency spectrum of the same signal. The effect of the non-linearity can be seen in the frequency spurs and in the increased noise floor (−25 dB to −30 dB). Thus it is desirable to reduce the effect of the non-linearity, ideally to less than the thermal noise of the GRO. 
         [0039]    A preferred embodiment of a GRO that yields improved resolution and reduced non-linearity with respect to the GRO embodiment described above is shown in  FIG. 12A . In this improved configuration of the GRO  100 , each delay stage has multiple weighted delay elements  42 . Connections to the multiple delay elements  42  are made to multiple delay stage outputs from around the GRO  100  with a connection matrix  104  in such a manner that the effective delay for each stage decreases as described without the gating functionality in S. S. Mohan et al., “Differential Ring Oscillators with Multipath Delay Stages,” IEEE 2005 Custom Integrated Circuits Conference, pp. 503-506, incorporated by reference herein in its entirety. 
         [0040]      FIG. 12B  illustrates a configuration for the delay stage used in the GRO  100  of  FIG. 12A  in which PMOS and NMOS transistors function as the delay elements  42 . In this configuration the number of delay elements  42 , the transistor sizes, and the connection matrix  104  are engineered to obtain an effective delay that is a factor of about 5-10 times smaller than the serial GRO  60  shown in  FIG. 7  and described above. 
         [0041]    The desired mode of oscillation, called the primary mode, is when the GRO  100  transitions occur in a serial sequence similar to the serial GRO  60 , albeit much closer together. However, due to the more complex structure of the multipath GRO  100 , modes of oscillation other than the primary mode can occur as well, and need to be suppressed for optimal operation of the GRO  100 . It has been found that a prime number N of delay stages helps to suppress other modes. In addition, the connection of delay elements to delay stage outputs can be approximately restricted to the preceding N/3 stages, with at least a minimal weighting of the elements to be distributed across these N/3 stages. 
         [0042]    The reduced propagation delay achieved using the illustrated configuration can result in a reduction of 10 dB or more in TDC error. The non-linearity resulting from the enabling and disabling of the GRO  100  is also reduced. In effect, many stages are in transition at any one time, therefore the non-linear time to enable and disable is less affected by a single voltage waveform, leading to an improvement of 10 dB or more. Quantization noise approaches the thermal noise, leading to more linear behavior in feedback loop applications. 
         [0043]    Due to the increased speed of the multipath GRO, the counting of transitions from each delay element  42  with the measurement module as described above for the serial GRO  60  and shown in  FIG. 7  may be difficult or inefficient. Alternatively, the state of a GRO operating in its primary mode can be found by counting the transitions of one output and sampling the entire oscillator output states. Using this configuration of the measurement module, the predictability of the GRO output state progression is relied upon to accurately map the observed oscillator state into measured time or phase. Because a larger number of GRO output stages can be in transition during the disabled measurement time, it is difficult to predict or establish a reliable GRO output stage transition pattern across the entire oscillator. To address this problem, the GRO output stages can be sub-divided and connected to appropriately sub-divided measurement module cells that each has a unique and predictable mapping from GRO state to time or phase. The full measurement module then sums together the measurement module cell outputs to result in the total time-to-digital converter output. 
         [0044]    A method for ensuring that transitions are not miscounted or corrupted during the measurement is an important part of achieving noise shaping, and was described above with reference to  FIG. 7 . Another method to remove glitches from the critical counter inputs is to utilize the monotonically increasing phases of the GRO. That is, the GRO phase can only increase, and therefore any transition glitches without significant motion of the GRO state can safely be ignored. Digital logic within the measurement module that crudely observes the GRO state can be combined with a transparent latch to accomplish this de-glitching. 
         [0045]    The invention also contemplates other GRO configurations in which variations on the serial or multipath GRO are utilized. In one such alternative configuration, multiple GROs are coupled to operate in sequence so that there is a fraction of a delay offset between each pair of GROs. The coupling can be accomplished using an appropriate connection matrix to achieve a finer resolution than is possible with a single GRO. Still other configurations of multiple ring oscillators used to achieve resolution improvement as are known in the art are contemplated for use in the GRO of the invention. 
         [0046]    While the invention has been shown and described with reference to specific embodiments, it should be understood by those skilled in the art that various changes in form and detail may be made therein without departing from the spirit and scope of the invention.