Abstract:
A radio frequency diode detector has a set of diodes having a differential voltage output, and a current source electrically coupled to the ring of diodes, the current source coupled to provide a forward bias current. This is followed by nonlinear signal processing to create an overall linear detector suitable for use in microwave power measurement.

Description:
BACKGROUND 
     Many situations require the detection of power in microwave communication links such as point-to-point communication links for high speed data communications. These wide band communication links run at high gigahertz frequencies, typically up to 24 GHz, and incur the need to measure carrier power in both transmitters and receivers over a wide dynamic range. In fact, the determination of power is indirect. Practically all integrated circuit (IC) solutions used in power measurement applications actually measure the voltage arising across a known load impedance, usually, but not necessarily, 50Ω. For example, a signal of amplitude 223.6 mVrms develops a power of P=V 2 /R=0.2236 2 /50=0.001 W. RF systems universally express power levels in milliwatts, and use the decibel equivalent expressed in dBm. The power developed by a 223.6 mVrms signal into a 50Ω load is 1 mW, which in decibel form is 0 dBm. 
     This power level is roughly at the center of typical ranges which need to be measured, ranging roughly from −30 dBm to +13 dBm. In voltage terms, this requires the determination of voltages spanning the range 7.07 mVrms to 1 Vrms, or about 141:1 in voltage terms and 43 dB in power terms. 
     In the past, RF power measurement has often employed logarithmic amplifiers (log amps) to address the dynamic range challenge. However, at microwave frequencies, suitable log amps require the use of the fastest available IC processes. They are also relatively complicated circuits, needing voltage references to establish their scaling rules, slope and intercept, and consume a great deal of supply current, typically on the order of 100 mA. 
     On the other hand, detectors based on Schottky diodes inherently have a rapid response, and have been used for decades in power measurement applications. They are now available in many IC processes. Their appeal lies in the fact that, once the RF amplitude is determined (usually referred to as the ‘envelope response’) the remaining signal processing can be carried out at relatively low frequencies, and typically at less than 1/100-th the supply current needed for a log amp. Further, the detection response can readily extend to signal frequencies of 100 GHz, roughly four times higher than the fastest available log amps. 
     Unfortunately, conventional diode detectors have an extremely nonlinear response. This is especially troublesome at small signal amplitudes; their output at low levels follows a roughly square-law characteristic, and, consequently, they are very insensitive to small signals. As usually implemented, they also have inherently poor temperature behavior, which erodes accuracy. These limitations are overcome by the measures described herein. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows a prior embodiment of a radio frequency (RF) diode detector. 
         FIG. 2  shows a graph of a prior art detector response. 
         FIG. 3  shows a typical embodiment of an RF detection system. 
         FIG. 4  shows a typical embodiment of an RF diode detector. 
         FIG. 5  shows an embodiment of a portion of an RF detector. 
         FIG. 6  shows an embodiment of a nonlinear function circuit. 
         FIG. 7  shows a more detailed embodiment of a nonlinear function circuit. 
         FIG. 8  shows an example of the improved detector&#39;s amplitude response. 
         FIG. 9  shows an embodiment of a converter circuit for an output of an RF diode detector. 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1  shows a prior art embodiment of a radio frequency (RF) power detector. Here, the RF generator  12  generates the signal power whose amplitude the detector must measure; the resistor  14  represents the source resistance of this generator. The diode  16  may consist of a Schottky diode having a metal-to-semiconductor interface, rather than a semiconductor-to-semiconductor interface, as in a traditional diode. A load capacitor  18  and load resistor  19  connect the output of the diode  16  to ground. The polarity direction of the diode is not of operational significance, but  FIG. 1  shows the RF signal applied to the anode of the diode  16 , with the output taken from the cathode. 
     In a simple analysis, it is stated that when the load capacitor  18  becomes fully charged, no current flows in the diode  16  and the circuit therefore becomes a ‘peak detector.’ In practice, this state of affairs is substantially altered by the presence of the load resistor  19 , which discharges the load capacitor  18  between the peak instants of the signal, to a greater or lesser degree. There is a finite average current in the diode  16 , causing the output voltage to be less than the peak voltage of the RF input. Because of the diode&#39;s voltage/current behavior, even small amounts of average current introduce a very significant voltage drop. This reduction in response becomes very pronounced at even moderate signal levels, and profoundly so at millivolt-level RF inputs. 
     This simple type of detector suffers from other problems in measurement applications, one of which is its extreme variation in scaling accuracy over its operating temperature range.  FIG. 2  shows an example of the amplitude response for this type of detector, demonstrating the loss of sensitivity at lower powers. The response curve  22  demonstrates the response at moderate temperatures. At low temperatures, the response would follow the curve  24 , while at higher temperatures, it would appear more like the curve  26 . 
     This variation across the operating temperature range raises a serious problem with this simple detector embodiment. A further problem arises because the input impedance of such a detector is also nonlinear, and is prone to generating even-order harmonic signals which are reflected back into the source. A practical transmitter or receiver channel is very susceptible to such distortion components. It should be noted here that modern systems have to meet extremely stringent specifications in this regard, and therefore it is evident that a detector that does not generate such distortion would be valuable. 
       FIG. 3  shows an embodiment of an RF detection system. The detector  40  provides input to a nonlinear function circuit  60 . The elaborated system may optionally include a post-detection RMS (root mean square) module  80  and a log amp module  100 . The RMS module  80  converts a fluctuating output of the detector to a steady state signal more closely representing the signal power, and the optional log amp module  100  may then convert this to a decibel representation. These modules may be used or not depending on the overall requirements of the measurement system. 
       FIG. 4  shows an embodiment of an improved diode detector having a high dynamic range and stable operation over temperature. The RF source  12  and its source resistance  14  are connected to a set of diodes  42 , which, even in the absence of an applied signal, are held in conduction by the small bias current I D . In this way, the diodes  42  are already operating in forward bias when the applied RF voltage is small, avoiding the insensitivity at the lower levels. The current flows through diode  41  to diode  43 , and simultaneously through diodes  45  and  47 . Two charge-holding load capacitors  46  and  49  are now used, one following positive signal excursions and the other following negative signal excursions. These connect to circuit nodes  33  and  35 , respectively, physically placed close to the input node  31 , making a very short RF path. 
     Due to the use of equal bias currents, there is no net DC voltage developed across the source resistance at the input node  31 , which is at ground or zero potential. Also, in the absence of an RF input, the output of diode  41  is likewise returned the ground potential, due to the matching of the equally sized diodes, having the same saturation current (herein denoted by I S ) as is familiar in semiconductor device theory. With the application of an RF signal, the voltage on the circuit node  33  increases in a positive direction with increases in the signal amplitude, and the voltage on the output node  102  follows this increase, starting from a zero-input baseline of zero. Similarly, with the application of an RF signal, the voltage on the circuit node  35  increases in a negative direction with increases in the signal amplitude, and the voltage on the output node  104  follows this voltage, again starting from a zero-input baseline of zero. 
     This circuit develops two voltage outputs, which, for large signal inputs essentially follow the positive and negative peaks of the applied RF signal. This provides a voltage-doubling response, without the voltage drops normally encountered in a simple diode detector. 
     Further, due to the symmetry of the  FIG. 4  circuit, the input impedance is much more linear than in prior art embodiments. In the prior art embodiments, when current flows only during on-signal polarity, even-order distortion components are reflected back into the source. In this embodiment, only weak odd-order reflections occurs. Typically the source would consist of power amplifier whose output drives an antenna, but a small sample of which is delivered to the detector by the directional coupler. With the asymmetric half-wave rectifier of the prior art, even-order distortion components may couple back to the antenna and negatively impact the transmitted signal. Similar problems arise in a receiver channel. Odd-order distortion is generally more tolerable in this context. 
     In a practical embodiment, the diodes are ‘doubled-up,’ using eight equally-sized devices, as shown. This is a layout measure taken to mitigate the offset-inducing effects of various chip gradients, such as temperature, strain, etc., and merely one embodiment, which may not be necessary or desirable, depending on the practical circumstances. 
     The circuit topology of  FIG. 4 , with the pre-biasing currents I D , may improve the detector response in the several ways just described. But the detector still responds in an essentially square-law manner to small signal amplitudes, becoming progressively more linear at higher input levels. Referring back to  FIG. 3 , the nonlinear function circuit  60  may allow the response to be linearized. 
       FIG. 5  shows a portion of the system of  FIG. 3 , focusing on the relationship between the detector and the nonlinear function circuit  60 . An appropriate example of a nonlinear function circuit  60  would be a ‘square rooting’ circuit that produces an output that is the square root of the input. In  FIG. 5 , the outputs of the detector from  FIG. 4  are first buffered. The output voltages of the detector are very small at the low end of the input amplitude range. For example, at an input of −35 dBm at 50Ω, the signal amplitude for a sine wave input is only 4 mV, and the corresponding output voltage of the detector cell will be in the microvolt range. 
     It is important that the bias currents I D  be very well matched. It will further be apparent that the loading of the outputs of this detector may be only very lightly loaded. This is the function of the buffer amplifier stages  51  and  53  in  FIG. 5 . These need not be unity-gain buffers, although, in view of the typically small supply voltage of a practical IC and the voltage doubling property of the detector, this may be the preferred choice. 
     The buffered outputs are applied to the nonlinear function circuit  60 , which is arranged to have the inverse amplitude response of the detector at low signal levels, which as we have seen is square-law. The appropriate nonlinear function is the square-root function. In theory, this function has infinite incremental gain at zero input. In practice, its implementation requires the provision of very high increment gain for small inputs. 
     It has been explained that while diode detectors exhibit a very weak response to small input signals, this response becomes asymptotic to a peak-detection, linear function at high input levels. One of the requirements of a fully-linearized RF voltmeter, which is a more accurate description of the function of integrated-circuit “power” detectors, is that the square-root function must be gradually “softened” or weakened at high signal levels. A practical implementation requires very careful characterization of the basic detector cell of  FIG. 4  to ensure an accurate overall fit to a fully-linear response. Further, the scaling of the square-rooting cell must be arranged to have the necessary temperature behavior. 
     The signal out from the detector, shown in  FIG. 4  as V OUT , becomes V IN  to the nonlinear function circuit  60 . This signal is also fed forward to the linear amplifier  55 . The nonlinear function circuit  60  injects a current I OUT  into the path of the linear amplifier  55  at the node where it connects to the input of the linear amplifier. Because there is a finite resistance at that node, the injected current causes the single-line output to be positive, where the output value can be viewed as flowing in the direction shown. Because the output current is the square root of the input voltage, eventually the nonlinear function circuit  60  will ‘turn off,’ or have no further effect on the output signal. The end result is that the output of the linear amplifier results from the nonlinear function circuit at low signal levels, and results from the detector response at higher signal levels (where the detector response becomes linear). 
       FIG. 6  shows one embodiment of the nonlinear function circuit  60 ; in this instance, a square root circuit. The current I 1  is connected to the transistor  54 , Q 2 . The reference current I R  is connected to the transistor  52 , Q 1 . The unknown current value I Z  is the one of interest and is connected to transistor  56 , Q 3 , and through it to the transistor  58 , Q 4 . The relationships are as follows:
 
 V   BE1   +V   BER   ≡V   BEZ   +V   BEZ ,
 
which leads to:
 
 V   K  Log( I   1   /I   S )+ V   K  Log( I   R   /I   S )≡ V   K  Log( I   Z   /I   S )+ V   K  Log( I   Z   /I   S );
 
where V K  is the thermal voltage of the transistors.
 
The thermal voltages cancel, as do the source currents I S . Using the rules of logarithms, the sum of Log I 1  and Log I R  results in Log I 1  I R  being equivalent to Log I Z  I Z . The logs cancel, and the result is that I Z  equals the square root of I 1  I R . This principle is what produces the inverse function of the detector. One should note that an emitter follower  63  connects to transistor  65  in a ‘tail chaser’ configuration for stability.
 
       FIG. 7  shows a more detailed embodiment of a nonlinear function circuit  60 . The current upon which the nonlinear function circuit  60  acts is a current proportional to the V IN  of  FIG. 5 . This current is I IN  for purposes of this discussion, but was earlier referred to as I OUT  in  FIG. 5 . The reference current I R  scales the whole function. The reference current comes from a reference generator that can be set with the desired scale factor. The amplifier  62  ensures that the voltage of the transistor  64  is sufficient to absorb the current I IN , and controls the reference point of the collector of the transistor  64 . Similarly, the amplifier  66  controls the reference point of the collector of the transistor  68 . 
     The summing node (NODE) of the output amplifier of  FIG. 5  has a voltage that is lower in voltage than the collector of transistor  70  can tolerate. One embodiment to rectify this situation replicates the current in transistor  72  to transistor  74 . Transistor  74  has a load resistor that takes the output (OUTPUT) almost to ground, and the NODE is set to a fixed voltage. One should note that the ground lines from resistor  78 , the transistor  64 , and the transistor  72  are typically tied together. This avoids any variation on the ground lines, in which even a millivolt variation can have negative effects on the operation of the circuit. The resistor  78  causes the output to behave in a ‘translinear’ way, in which it functions as a square root or other nonlinear function until the output of the linear path dominates the output of the circuit. 
     A translinear circuit is a circuit that functions under the translinear principle. The translinear principle states that in a closed loop containing an even number of translinear elements with an equal number of them arranged clockwise and counter-clockwise, the product of the currents through the clockwise elements equals the product of the currents through the counter-clockwise elements. In  FIG. 7 , the translinear loop is formed within the transistors  68 ,  74 ,  72 , and  65 . Under the translinear principle, similar to the discussion of  FIG. 6 , the product of the currents I W I W  in the clockwise loop equals the products of I TN  I R . The result is that I W  equals the square root of I IN  I R . I W′  as the output current is then equal to (A 1 /A 2 ) I W . 
       FIG. 8  shows a curve of a typical detector response. The detector response has a first region  67  often referred to as the ‘square law’ region, which is exponential in nature. The second region  69  is a linear region. A translinear circuit often operates to linearize the square law region of the response, and then letting the response remain linear in the linear region. The embodiments of the diode RF detector here employ this principle. 
     As mentioned previously, the nonlinear function circuit  60  operates on the input current I IN , which is proportional to the output voltage of the detector.  FIG. 8  shows an embodiment of a circuit that takes the output voltage and generates the proportional current. The circuit in this embodiment consists of a degenerated pair of transistors  82  and  84  having bias currents I E . They connect to the differential output voltages from the amplifiers  51  and  53  from  FIG. 5 . The voltage swing could have a wide dynamic range from 3.2 V (+15 dBm) to the millivolt range. The pair is connected to a current mirror  90  with some adjustments for offsets at OFFSET, allowing an adjustment at the bottom end of the range to ensure proper function of the circuit at the bottom of the range. 
     One should note that the pair of transistors  82  and  84  will more than likely not match perfectly, so they may be connected in a ‘cross-quad’ manner using transistors  86  and  88  to ensure stability of the transistors. Similarly, other transistors in the other figures and embodiments may also have cross-quad transistors, but these embodiments were not shown for simplicity. However, no limitation to using or not using cross-quad transistors is intended nor should any be implied. 
     In this manner, an RF power detector using diode detectors is created. It has high dynamic range, even at very high bandwidths in the hundreds of Gigahertz range, good temperature behavior, and high sensitivity. Different embodiments and implementations of specific components of the embodiments are shown here and any and all variations of these components and their configurations may be used in any and all of the different embodiments. 
     The embodiments described herein can be modified in arrangement and detail without departing from the inventive concepts. For example, some of the principles have been described above in the context of a radio frequency power detector using diodes. Nonetheless, the principles also have utility separately, and in other applications. Many numerical examples have been provided to facilitate explanation of the inventive principles, but the inventive principles are not limited to any such specific examples. Accordingly, such changes and modifications are considered to fall within the scope of the following claims.