Abstract:
Concurrently measuring, correlating, and processing magnetic and electric field data includes measuring base band signals, and then up-converting those band signals to a higher frequency for filtering, while at the same time preserving phase and amplitude information. All timed elements in the system are rigorously synchronized. The increased data set results in improved signal-to-noise ratio and information correlation.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This patent application claims the benefits of provisional patent application Ser. No. 61/362,241, filed Jul. 7, 2010, and provisional patent application Ser. No. 61/366,916, filed Jul. 22, 2010. 
     
    
     STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT 
       [0002]    None. 
       THE NAMES OF THE PARTIES TO A JOINT RESEARCH AGREEMENT 
       [0003]    None. 
       INCORPORATION-BY-REFERENCE OF MATERIAL SUBMITTED ON A COMPACT DISC 
       [0004]    None. 
       BACKGROUND OF THE INVENTION 
       [0005]    (1) Field of the Invention 
         [0006]    The invention relates to devices and processes for geophysical prospecting, and, more particularly, to the removal of noise typically associated with the data collection of Control Source Electromagnetic (“CSEM”) and Magnetoturelic (“MT”) signals. 
         [0007]    (2) Description of Related Art (Including Information Disclosed Under 37 CFR 1.97 and 1.98) 
         [0008]    There are many U.S. patents and patent applications related to electromagnetic surveying. Some of the more relevant ones appear to be the following: U.S. Pat. No. 6,253,100, for broad band electromagnetic holographic imaging; U.S. Pat. No. 7,203,599, for acquiring transient electromagnetic survey data; U.S. Pat. No. 7,337,064, for electromagnetic surveying for hydrocarbon reservoirs; U.S. Pat. No. 7,483,792, for electromagnetic surveying for hydrocarbon reservoirs; U.S. Pat. No. 7,502,690, for using time-distance characteristics in acquisition of t-CSEM data; U.S. Pat. No. 7,565,245, for electromagnetic surveying; U.S. Pat. No. 7,805,249, for controlled source electromagnetic surveying with multiple transmitters; U.S. Pat. No. 7,822,562, for removing air wave noise from electromagnetic survey data; U.S. Pat. No. 7,941,273, for using time-distance characteristics in acquisition of T-CSEM data; 20080105425, for electromagnetic surveying for hydrocarbon reservoirs; 20090005994, for time lapse analysis with electromagnetic data; 20090005997, for spatial filtering of electromagnetic survey data; 20090067546, for compensating electromagnetic data; 20090072831, for real time monitoring of the waveform transmitted by an electromagnetic survey; 20090082970, for electromagnetic surveying; 20090103395, for wavelet denoising of controlled source electromagnetic survey data; 20090120636, for controlled source electomagnetic surveying with multiple transmitters; 20090126939, for electromagnetic data processing system; 20090204330, for using time-distance characteristics in acquisition of T-CSEM data; 20090265111, for signal processing of marine electromagnetic signals; 20090276189, for estimating noise at one frequency by sampling noise at other frequencies; 20100018719, for inversion of CSEM data with measurement system signature suppression; 20100065266, for controlled source electromagnetic reconnaissance surveying; 20100176791, for correcting the phase of electromagnetic data; 20100224362, for electromagnetic imaging by four dimensional parallel computing; 20100233955, for electromagnetic air-wave suppression by active cancellation; 20110013481, for detecting marine deposits; and 20110087435, for electromagnetic prospecting waveform design. All of these patents and patent applications are incorporated herein by this reference. 
         [0009]    Several techniques exist that attempt to remove air wave noise and other noise sources from the signal of interest in a CSEM system. These techniques include active filtering, signal encoding such as grey coding, and noise estimation and subtraction at different frequencies. Additionally, until recently, all of these techniques were supplemented by physical isolation of the receiving elements from the noise source, by submersion in a marine environment, thus using the water as an air wave signal filter. These methods are typified in the above-listed U.S. patent applications 2009/0204330, 2009/0265111, 2009/0276189, 2011/0013481, and in U.S. Pat. No. 7,822,562. 
         [0010]    The major problem with these techniques is that they are unable to successfully filter out in-frequency noise because the frequency of interest is very close to the frequency of the noise, that is, typically between fifty and sixty hertz (50-60 hz). 
         [0011]    In addition, for the purposes of operation on the surface, there are many more sources of noise and amplification of noise, such as rail lines, pipelines and barbed wire fences, that is, anything that is ferrous and long. The typical solution to these noise problems is to survey the area before performing a CSEM survey, and remove the known anomalies from the data. Items can be missed in the preliminary survey, causing additional unexpected noise in the data, and thus reducing delineation and depth of investigation. A person skilled in the art of performing CSEM surveying will understand the issues that uncontrolled noise can cause when using existing systems for surface based measurements. The use of CSEM for surveying is described in U.S. Pat. No. 7,203,599. 
         [0012]    In light of the foregoing, a need remains for a system and method of visualizing sub-surface formations that reduces noise, and improves resolution. 
       BRIEF SUMMARY OF THE INVENTION 
       [0013]    The present invention improves the visualization of sub-surface formations in a static state by reducing noise, and improving resolution. Multiple simultaneous channels of E and H field data using high speed data acquisition techniques coupled with advanced noise filtering techniques and more precise determination of phase data, allows for the rapid interpretation of 2D, 3D, and 4D data in CSEM operations to greater depths and finer bin resolution. 
         [0014]    The receiver system of the present invention is able to detect the transient states being caused by either removing fluid or gas from the formation, or imposing fluid and propant under pressure during fracturing operations. 
         [0015]    The invention takes the differential signal and up-converts it to a higher frequency, imposes RF noise filtering techniques at the higher frequency, and preserves both phase and amplitude information from the original signal. The inventive technique allows software to control the frequency at which the system will collect data, and the frequencies of data that are rejected. The method of the present invention, includes a source clock with a low phase jitter. In addition, the current invention implements an enhanced method for obtaining induced magnetic field data that produces improved granularity in formation data. 
     
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
         [0016]      FIG. 1A  depicts a receiver layout in the form of a matrix of receivers. 
           [0017]      FIG. 1B  depicts an alternate receiver layout in the form of a string of receivers. 
           [0018]      FIG. 2  Is a block diagram of the modules that are contained within a receiver system. 
           [0019]      FIG. 3  is a block diagram of the electric field portion of the receiver system. 
           [0020]      FIG. 4  is a block diagram of the magnetic field portion of the receiver system. 
           [0021]      FIG. 5  is a block diagram of the timer module of the receiver system. 
           [0022]      FIG. 6  is a block diagram of the control module of the receiver system. 
           [0023]      FIG. 7  is a flow chart of the logic of the method of the present invention. 
       
    
    
       [0024]    In the figures, the left-most significant digit(s) in the reference numerals denote(s) the first figure in which the respective reference numerals appear. 
       DETAILED DESCRIPTION OF THE INVENTION 
       [0025]    In the preferred embodiment a plurality of receivers are arranged to collect data that is used to create images of the physical features within sub-surface formations. The receiver system measures the potential difference of the decaying electric field signal and surface currents caused by a CSEM transmitter pulse, between at least two widely spaced electrodes that are driven into the ground. In addition the receiver measures the magnetic fields that result from the excitation caused by a CSEM transmitter. The transmitter is a separate device. An example of a CSEM transmitter is a model no. DIT TX50 manufactured and sold by Deep Imaging Technologies, Inc., head-quartered in Houston Tex. 
         [0026]    Referring to  FIG. 1A , in the preferred embodiment a plurality of time-synchronized receiver systems  100  are assembled as depicted,  100   a,    100   b,    100   c,    100   d,    100   e,    100   f,    100   g,    100   h  and  100   i  to form a receiver matrix  101  around or offset from a wellhead  103 . In a typical setup the receiver matrix  101  extends over an area of one kilometer square  106 . Any number of receiver systems  100  can be used to form the matrix  101 . A CSEM transmitter  102  is placed in accordance with the requirements of the CSEM survey for a plurality of subsurface formations  104 , and is a distance of greater than five hundred meters  107 , from the middle of the receiver matrix  101  to ensure the CSEM transmitter  102  is offset from the receiver matrix  101 . The requirements for a CSEM survey can be understood by any person familiar with the practice of CSEM surveying. 
         [0027]    Referring now to  FIG. 1B , an alternate embodiment allows for the assembly of a plurality of time-synchronized receiver systems  100  to form a receiver string  105  as depicted by  100   a,    100   b  and  100   c.  In a typical setup the receiver string  105  extends over a distance of 1 KM  108  and the CSEM transmitter  102  is placed at a distance of approximately 500 meters  109 , from the receiver string  105  In a further alternate embodiment at least one receiver string  105  is used in conjunction with the action of repositioning the receiver matrix  100  after each set of data has been received and collected. All of the embodiments of the method of the present invention include the step of accurately locating the receiver systems  100  relative to each other and the CSEM transmitter  102 . That step is shown as step  725  in  FIG. 7 . 
         [0028]    Referring now to  FIG. 2 , the receiver system  100  includes several sub-systems and sensor groups. In the preferred embodiment, a dipole receiver  200  contains electrodes  201  and  202 . A magnetic field loop antenna  205  connects to an electronics assembly  210 . The electronics assembly  210  includes an electric field input filter card  215 , a surface electric current detector circuit  220 , a two-channel Software Defined Receiver (SDR)  225 , a magnetic field input filter card  230 , a magnetic point potential circuit  235 , a Magnetometer Card (MC)  240 , a Timing Module (TM)  245 , a Control, Digital Storage, and Communications Module.  250 , a GPS Module  255  and a Power Module  260 . 
         [0029]    Referring now to  FIG. 3 , in the preferred embodiment, for each of the receiver systems  100  the two electrodes  201  and  202  are widely spaced and inserted into the surface of the earth. The dipole receiver  200  receives the electric field resulting from a synchronized transmitter pulse. The electrodes  201  and  202  are conductively coupled to a pair of signal inputs  300   a  and  300   b  in the electronics assembly  210 . A pair of signals, ES 1  and ES 2  are received through the electrodes  201  and  202 , and pass into the E field Input Filter card  215 . A common mode amplifier  305  receives the signals ES 1  and ES 2  and outputs a difference signal ES 3  at an output  310 . The output  310  is conductively coupled to a power line filter  315 . The power line filter  315  can be optionally removed from the circuit through a user activated switch  302 , which is connected to a relay switch  320 . The output of the power line filter  315  is conductively coupled to an input channel of the SDR  225 . Alternatively, the output  310  of the common mode amplifier  305  bypasses the power line filter  315 , and conductively couples to a channel of the SDR  225 . 
         [0030]    The output  310  is conductively coupled to the surface electric current detector circuit  220 , and thus the ES 3  signal at the output  310  is passed through a low pass filter  322  and an amplifier  324 . An output signal ES 6 , at an output  325 , is the surface electric direct current (DC). The output  325  is conductively coupled to a 24-bit analog-to-digital converter (ADC)  600  shown in  FIG. 6 . 
         [0031]    The difference signal ES 3  is present at an input  323  (also known as “Port A” of the actual mixer circuit  330 ) to a channel  225   a  of the SDR  225 . The difference signal ES 3  is passed to a capacitively coupled input  327  of a high dynamic range mixer circuit  330 . In the preferred embodiment, the mixer circuit  330  is a MiniCircuits Model SBL- 1 A+ (DC-100 MHz version), manufactured by MiniCircuits in Brooklyn, N.Y. 
         [0032]    Input  326  (also known as “Port B” of the actual MiniCircuits mixer circuit  330 ) is supplied with a 25 dbm signal  515   a,  generated from a high stability source in the timing module  245 . The frequency of the signal  515   a  is controlled by software executing in a micro processor in the Control, Digital Storage, and Communications Module  250 . The frequency of the signal  515   a  can be set to any one of a wide range of frequencies. In the preferred embodiment of the present invention the frequency is set to 9 Mhz. 
         [0033]    The frequency of the incoming signal ES 3  is up-converted by the mixer circuit  330 . An output of the mixer circuit  330  is connected conductively to a 40 db gain IF amplifier  335 . An output  340  of the IF amplifier  335  is conductively connected to the input of a combination Cohn filter and a diplexer circuit  345 . The diplexer circuit is to help with matching and minimal phase distortion. An example of a Cohn filter design is one sold by Clifton Laboratories, Clifton, Va., and can be understood by a person familiar with the art of RF filter design. In an alternate embodiment, the Cohn filter in the Cohn filter and diplexer circuit  345  can be replaced by any RF band pass filter that can be digitally controlled. 
         [0034]    The output of the Cohn filter and the diplexer circuit  345  has a signal ES 5 . The output of the Cohn filter and the diplexer circuit  345  is conductively coupled to a low noise Intermediate Frequency (IF) amplifier  350 . In the preferred embodiment, the amplifier  350  is the Analog Devices AD9855, manufactured by Analog Device Inc, Norwood, Mass. 02062-9106. The amplification of the signal ES 5 , in the IF amplifier  350  stage is 12 db, and is designed to regain the signal loss through the Cohn filter and the diplexer circuit  345 . 
         [0035]    The output of the IF amplifier  350  is conductively coupled to the input of an Enhanced Tayloe detector (ETD) circuit  355 . At the input to the ETD circuit  355  the signal is coupled to a power splitter  355   a.  The power splitter  355   a  galvanically isolates two outputs  355   b  and  355   c  that are conductively coupled to two separate tayloe detector circuits  355   d  and  355   e  respectively. The tayloe detector circuits  355   d  and  355   e  are implemented using CMOS components. Using two separate tayloe detector circuits  355   d  and  355   e,  for resolving the in-phase and quadrature signals from the same input signal ES 5 , reduces cross talk and signal noise between two outputs  360  and  365 . 
         [0036]    The tayloe detector circuits  355   d  and  355   e  are fed two 8,999,000 Hz clock signals, an in-phase clock signal  515   a,  and a quadrature out-of-phase clock signal  515   b  respectively. (Signals  515   a  and  515   b  are shown only in  FIG. 5 .) The signals  515   a  and  515   b  are generated from the same source in the timing module  600  (shown in  FIG. 6 ), and are separated by a phase shift of 90 degrees. The tayloe detector circuits  355   d  and  355   e  generate an in-phase signal Ei present at  355   f  and a quadrature signal Eq present at  355   g  respectively, at frequencies between 0.01 Hz and 50 KHz. The in-phase signal  355   f  contains the amplitude information, and the quadrature signal  355   g  contains the phase information, of the original electromagnetic field signal ES 3  at the output  310 . 
         [0037]    The Enhanced Tayloe detector circuit  355  can be understood by any person skilled in the art of superheterodyne radio frequency (RF) design. 
         [0038]    The signals Ei, present at  355   f,  and Eq, present at  355   g,  are passed to bandpass diplexer networks  355   h  and  355   i  respectively. The outputs  360  and  365  of the bandpass diplexer networks are each buffered by low noise amplifiers  362  and  367  respectively. The low noise amplifiers  362  and  367  output a pair of signals  370  and  375  respectively that are each passed to the control module  250 . The bandpass diplexer networks  355   h  and  355   i  can be understood by a person familiar with the art of RF radio design. An example of a diplexer design is at the Amateur and Short Wave Radio Electronics Experimenter&#39;s Web Site. 
         [0039]    The three streams of digital data representing the instantaneous values of the signals Ei, present at  370 , Eq, present at  375 , and ES 6 , present at  325 , over time are stored in the control module  250 , in a bulk memory store  615 . The data is stored in the industry standard SEG-D format. 
         [0040]    Referring now to  FIG. 4 , in the preferred embodiment the H-field detector consists of the loop antenna  205 , the filter card  230 , the magnetic point potential circuit  235 , a second channel  225   b  of the SDR  225 , and the magnetometer card  240 . 
         [0041]    The signals HS 1  and HS 2 , from the magnetic field loop antenna  205 , are present at inputs  400   a  and  400   b.  The inputs  400   a  and  400   b  are coupled to an H field filter card  230 . The output of the H field filter card  230  is conductively coupled to channel  255   b  of the SDR  225 . A second channel  225   b  of the SDR  225  has two output signals, Hi present at  470 , and Hq present at  475 , that are passed through to the control module  250 . 
         [0042]    The assembly, purpose, and operation of the circuit elements and sub elements within the depicted blocks  230 ,  225   b,  an Enhanced Tayloe detector (ETD) circuit  455 , and  235 , and the depicted sub elements  400   a,    400   b,    405 ,  410 ,  415 ,  420 ,  422 ,  424 ,  426 ,  430 ,  435 ,  440 ,  445 ,  450 ,  455   a,    455   b,    455   c,    455   d,    455   e,    455   f,    455   g,    455   h,    455   i,    460 ,  465 ,  462 ,  467 ,  470 ,  423 ,  427  and  425  in  FIG. 4 , are identical to those of the elements  215 ,  225   a,    335  and  220 , and the depicted sub elements  300   a,    300   b,    305 ,  310 ,  315 ,  320 ,  323 ,  327 ,  326 ,  330 ,  335 ,  340 ,  345 ,  350 ,  355   a,    355   b,    355   c,    355   d,    355   e,    355   f,    355   g,    355   h,    355   i,    360 ,  365 ,  362 ,  367 ,  270 ,  375 ,  322 ,  324  and  325  respectively in  FIG. 3 , as recited in the description for  FIG. 3 . 
         [0043]    In an alternate embodiment the wire loop in the loop antenna  205  is replaced by a solenoid. The solenoid is a wire wound core with a high number of turns of Linz wire, and is center tapped. 
         [0044]    The signal  620   a  (see  FIG. 6 ) is passed to the buffer  460  and to the field null coil  487 . The signal  620   a  can be derived from a potentiometer or the output of a Digital to Analog converter  620  (see  FIG. 6 ) and is used to null the local magnetic field from the magnetometer 
         [0045]    At least one of the outputs on an X-axis magnetometer  485  can be coupled to at least one of the three inputs in the magnetometer card  240 . Local ambient field effects are negated by a field coil  487 , as is typical in CSEM systems. In the preferred embodiment the X-axis magnetometer  485  is designed to detect fields that are parallel to the earth&#39;s surface. At least one of the magnetometer outputs is coupled to a divide-by-n counter  495 . The output of the divide-by-n counter  495  is passed to a capture-and-compare input in a microprocessor in the control module  250 . 
         [0046]    The four signals, Hi present at  470 , Hq present at  475 , HS 6  present at  425 , and Mx 1  present at  498 , are passed to the control module  250 . The four streams of digital data representing the instantaneous values of the signals Hi, Hq, Hs 4 , and Mx 1 , over time are stored to the control module  250 , in a bulk memory store  615 . The data is stored in the industry standard SEG-D format. 
         [0047]    Referring to  FIG. 5 , the timing module  245  receives at least one low drift, phase accurate timing signal. In the preferred embodiment the master clock is sourced from a 400 Mhz oscillator  500 . In the preferred embodiment, the oscillator is the NBXSBB023 400 Mhz LVPECL clock oscillator selected for 20 ppm accuracy, manufactured by On Semiconductor of Phoenix, Ariz. 
         [0048]    The 400 Mhz oscillator  500  is connected a Complex Programmable Logic Device (CPLD)  505 . The signal is divided down to a 50 Mhz clock signal  505   a  and a 27 Mhz clock signal  505   b.  In the preferred embodiment, the CPLD  505  is the Xilinx 3C256 CPLD, manufactured by Xilinx, Inc. 2100 Logic Drive, San Jose, Calif. 95124 U.S.A. The CPLD  505  is partially programmed as a divider, and is controlled by the control module  250  through a control bus  605   a  (shown in  FIG. 6 ). The 50 Mhz clock signal  505   a  is coupled to a Direct Digital Synthesis (DDS) device  515 , the semiconductor AD 9958 manufactured by Analog Devices Inc of Norwood Mass., USA. The DDS device  515  is used to create two clock signals. The first clock signal is in phase with the 400 Mhz oscillator  500 , and is the in-phase clock signal  515   a.  The second clock signal is offset by 90 degrees in phase from the 400 Mhz oscillator  500  and is the out-of-phase clock signal  515   b  . The in-phase clock signal  515   a  and the out-of-phase clock signal  515   b  are fed to the Enhanced Tayloe circuits  355  and  455 . In addition the in-phase clock signal  515   a  is fed to the mixer circuits  330  and  430 . The 27 Mhz clock signal  505   a  is supplied to the ADC  600  (shown in  FIG. 6 ). 
         [0049]    The CPLD  505  is synchronized to other receiver systems  100 , through a synchronization pulse  525   a  from a GPS module  525 . An exemplary piece of equipment to perform receiver system location and synchronization is a PG11 Global Positioning System receiver, manufactured by Laipac Tech of Richmond Ontario Canada. Synchronization of the CPLD  505  using the synchronization pulse for the GPS module  525 , coupled with compensation for distance to satellite delays, provides for a method to completely synchronize all the receivers and transmitters in a CSEM setup. 
         [0050]    In addition, a GPS serial data stream  525   b  is passed to the control module  250  for storage of location information. 
         [0051]    In an alternate embodiment the 400 Mhz oscillator circuit  500  is input into a low jitter, low phase noise clock distribution semiconductor (CDS). The CDS generates the 27 Mhz clock  505   a,  for the ADC  600  and the 50 Mhz clock  505   b  for the DDS device  515 . An example of a CDS is the AD 9521, manufactured by Analog Device Inc, Norwood Minn. U.S.A. 
         [0052]    In another alternate embodiment the master clock is a rubidium atomic clock. In another alternate embodiment synchronization can also be achieved through a timing module  605 , shown in  FIG. 6 . 
         [0053]    Referring to  FIG. 6 , the control module  250  receives a plurality of signals  602  from the E surface electric current detector circuit  220 , the magnetic point potential circuit  235 , and the two channel SDR  225 . The signals  602  pass through a plurality of clipper circuits  625  that are used to limit the amplitude of the input to the ADC  600 . The ADC  600  can sample the incoming signals at any rate from  3 . 0  K samples per second (sps) to 255 K sps. The ADC  600  allows for significant oversampling of the data stream. In the preferred embodiment, the ADC  600  is the AD 1278 manufactured by Analog Device Inc, Norwood, Minn. 
         [0054]    The clipper circuits  625  are synchronized by a signal  630  from the microprocessor  605  that uses data from the magnetometer card  240  to detect the air wave. The signal from the magnetometer card  240  causes the clipper circuit  625  to attenuate the received signals  602  until the airwave has passed. The signals  602  pass into the ADC  600  and are converted to a digital data stream that is passed to a microprocessor  605 . In the preferred embodiment, the microcontroller  605  is the AVR32 manufactured by Atmel of San Jose, Calif. 
         [0055]    The microprocessor  605  moves the data stream from the ADC  600  and stores the data stream in a bulk memory  615 . The microprocessor also receives location information from the GPS serial data stream  525   b  and stores the data in the bulk memory  615 . 
         [0056]    A communications module  610  connects to a user interface (UI)  625 . The UI  625  can be used to adjust and control aspects of the operation of the receiver system  100 . In the preferred embodiment the UI  625  consists of a display and a user input device. In an alternate embodiment of the UI  625 , the input is achieved through a series of switches and potentiometers. 
         [0057]    A Digital to Analog Converter (DAC)  620  outputs a signal that is varied under microprocessor control until the ambient magnetic field is nulled in the X axis magnetometer  485 . 
         [0058]    Referring to  FIG. 7 , a software control program  700  executes on the microprocessor  605 . The software control program  700  consists of a series of steps that can be controlled or adjusted by input from the user interface (UI)  625 . The software control program  700  reads the input settings in step  705  from the front panel of the UI  625 , and saves the settings to the bulk memory  615 . The input settings define the SDR  225  by setting the up-converter frequency for the in-phase clock signal  515   a,  the down converter frequency for the in-phase clock signal  515   a,  and quadrature clock signal  515   b  of the enhanced Tayloe detectors  355  and  455 . The combination of setting the up-converter frequency and the down converter frequencies creates an accurate software-defined band pass filter with a software-controlled center frequency. 
         [0059]    In step  710 , the software control program  700  initializes the flash file. Step  715  normalizes the magnetometer  485  to the linear region of operation. Step  720  waits for the air wave to be detected. Once the air wave has been detected, step  725  starts the DDS device  515  and CPLD  505  at a precise start time using the GPS Module  525  synchronization pulse  525   a.    
         [0060]    In step  730  the ADC  600  data rate is set initially to 190 Khz, and the ADC  600  is started. In step  735  the processor reads the ADC  600  data on an end of conversion interrupt from the ADC  600 . In step  740  the data is stored in a standard format to the bulk memory. In step  745  the user interface display is updated, and any required data transmission is done through the communications module  610 . 
         [0061]    The system returns to step  735  to await the next end of conversion interrupt from the ADC  600 . The system continues in a loop  750  until all data has been collected and stored into bulk memory  615 . In an alternate embodiment, the step  705  includes the addition of entering, via the UI  625 , a pre-defined range of frequencies that the SDR  225  will sweep through during data collection. 
         [0062]    In operation, a plurality of receiver systems  100  are arranged as depicted in either  FIG. 1A  or  FIG. 1B . The receiver systems  100  are all synchronized through the synchronization pulse generated by the GPS module  225  in each receiver. In addition, location information is stored from the GPS module  225 , along with the SEG D data saved in the bulk memory  615 . The CSEM transmitter  102  is also capable of synchronization from a GPS synchronization pulse. 
         [0063]    Once a transmitted wave has been generated, each receiver detects the air wave, and attenuates the data to the ADC  600  present in each receiver system  100 . Once the air wave has passed, each receiver system begins collecting data. 
         [0064]    The E field dipole receiver  200  is designed to detect changes in the electric field created by an active transmitter pulse or passively from spontaneous potentials. 
         [0065]    The H field loop antenna  205  is designed to detect changes in the ambient magnetic field in all orientations, except those parallel to the earth&#39;s surface, caused by induced eddy currents in underground formations. The eddy currents induce magnetic fields that are of short duration. The loop antenna  205  is of typical design for this application, and varies in diameter dependent on depth of investigation required. The diameter can exceed  150  meters. The magnetometer card  240  provides the last axis of information that is combined with the magnetic field loop antenna  205  axis information, to determine the source of the arriving magnetic waves. 
         [0066]    The received difference signals ES 3  and HS 3  are processed as depicted in  FIGS. 3 and 4  respectively. Referring to  FIG. 3 , the combination of the mixer circuit  330 , Cohn filter and diplexer circuit  345 , and the enhanced Tayloe detector circuit  355 , result in phase coherent, noise-free data. 
         [0067]    In operation the Cohn filter and diplexeor circuit  345  is used in the band pass mode, and software executing on the microprocessor  605  controls the center frequency of the bandpass filter by controlling the up-converter frequency in the mixer circuit  345 . The skirts of the Cohn filter and diplexer circuit  345  are very tightly defined and drop off at better than −70 db per decade, and allow the SDR  225  to provide a bandpass that can be set to different frequencies of interest. The frequency content of the incoming signal ES 3  present at  310  is reduced to the range of interest at the up-converted frequency. The Cohn filter and diplexer circuit  345  is also known as a minimum loss filter, and has very high Q factors, in excess of 10,000. 
         [0068]    The benefit of using this technique to filter out unwanted signals from ES 3  and HS 3  can be understood by any person skilled in the art of superheterodyne-based Software Defined Receivers. In addition, this method brings added benefit to the post-processing of data, because the frequency of the recorded data is tightly defined, and provides additional constraints for data processing. 
         [0069]    The signal that results from the Cohn filter and diplexer circuit  345  has low noise content and low phase shift. The signal passes through the enhanced Tayloe detector  355 , is down converted as a result of the function of the Tayloe detectors, and is split into amplitude (in-phase signal) and phase (quadrature signal) components, again with low noise content and good phase accuracy. 
         [0070]    The data in the form of amplitude and phase pairs for each of the electric and magnetic fields is passed to the ADC  600 . 
         [0071]    It is an important element of the current invention that each channel of data being processed by the ADC  600  in the control module  250  has its sample start time synchronized in the pico second, or shorter, time frame. In addition, due to conductor line latencies, the time-critical ADC conversions all occur on a single chip, the AD 1278, and are concurrent to within 50 pico seconds. There are other latencies in the system, partly due to cable length variations and other factors that must also be measured. This is done by applying a test signal from the transmitter that is synchronized with the receivers, using a precise clock, and monitoring for arrival times at each of the ADC  600  inputs, and synchronized against a precise clock. This procedure must be performed for all receivers in the system and a calibration factor is programmed for each channel in the factory. 
         [0072]    The system collects amplitude, phase and point potential data from the magnetic (H) field that is stored in the bulk memory  615 . The phase data storage of the H field is unique to the current invention. 
         [0073]    The present invention benefits from the use of the enhanced Tayloe detector circuit  225  in each of electric and magnetic receiver channels, and the high speed ADC  600 , because the recorded data has a low signal-to-noise ratio, better than −120 dbm, and low phase distortion, less than 0.01%. The data stored in the bulk memory  615  includes additional data that creates a rich data set. The additional data items are a magnetic quadrature output signal Hq present at  475 , a surface current signal ES 6  at output  325 , and a magnetic point potential signal HS 6  at output  425 . The data are all synchronized to the system clock, or to an atomic clock, or to a GPS synchronization pulse. The precise timing of all ADC acquisition cycles allows for improved resolution at sub-surface depths, beyond 10,000 meters. In addition, reduced “bin” size is achieved. 
         [0074]    A “bin” in this context is a location of a finite size, usually a cube, within a mathematical representation (2D, 3D, or 4D array) of sub-surface geology. The bin is used to accumulate some predetermined value or combination of values for the location in the sub-surface geology. 
         [0075]    The preceding is merely a detailed description of one (or more) embodiments of the invention. Numerous changes to the disclosed embodiments can be made in accordance with the disclosure herein, without departing from the spirit or scope of the invention. The preceding description, therefore, is not meant to limit the scope of the invention. Rather, the scope of the invention is to be determined by only the appended claims and their equivalents.