Abstract:
An LED (light-emitting diode) driver for a photoplethysmography system, including a switched-mode operational amplifier for driving a driver transistor with a source-drain path in series with the LED. In a first clock phase in which the LED is disconnected from the driver transistor, the amplifier is coupled in unity gain mode, and a sampling capacitor stores a voltage corresponding to the offset and flicker noise of the amplifier; the gate of the driver transistor is precharged to a reference voltage in this first clock phase. In a second clock phase, the sampled voltage at the capacitor is subtracted from the reference voltage applied to the amplifier input, so that the LED drive is adjusted according to the sampled noise. A signal from the transmitter channel is forwarded to a noise/ripple remover in the receiving channel, to remove transmitter noise from the received signal.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This application claims priority, under 35 U.S.C. §119(e), of Provisional Application No. 62/236,589, filed Oct. 2, 2015, incorporated herein by this reference. 
     
    
     STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT 
       [0002]    Not applicable. 
       BACKGROUND OF THE INVENTION 
       [0003]    This invention is in the field of circuitry for driving light-emitting diodes (LEDs). Embodiments are more specifically directed to LED driver circuitry in transmitters of photoplethysmography (PPG) systems. 
         [0004]    Photoplethysmography (PPG) is a modern technology that has proven useful for the measurement of cardiovascular function in humans. According to this technology, fixed wavelength light from a light-emitting diode (LED) is emitted into the skin of a human subject, and is sensed by a photodiode (PD) after transmission through the skin and underlying tissue. The characteristics of the sensed light allows measurement of medical parameters such as oxygenation, pulse rate, respiratory function, and the like. 
         [0005]    Conventional PPG sensors include the well-known pulse oximeter, for example of the type that clip-on onto the finger of the patient. Pulse oximeters typically measure the oxygen saturation of circulating blood from a comparison of the absorption of light in the dermis and subcutaneous tissue at two different wavelengths. So-called “wearable” devices such as heart rate sensors also utilize this technology, but need only measure light absorption at a single wavelength. 
         [0006]      FIG. 1  generically illustrates the architecture of a conventional PPG system. Transmitter  3  in this system includes LED  2 , which has its anode biased by the V dd  power supply voltage and its cathode coupled to ground via LED driver  4 . As mentioned above, PPG pulse oximeters will included multiple LEDs  2  (e.g., red and infrared), operated in time-multiplexed fashion. When forward-biased by LED driver  4 , LED  2  emits light into the patient, for example the index finger of the patient. Receiver  7  includes photodiode  6 , which has its cathode biased at the V dd  power supply voltage and its anode connected to the input of amplifier  8 , and which is normally reverse-biased so that photons impinging photodiode  6  will produce a current detectable by amplifier  8 . In this manner, photodiode  6  senses the extent to which the light emitted by LED  2  is transmitted through the subject. The output of amplifier  8  is forwarded to the desired processing and analysis circuitry of the PPG system to determine the desired medical measurement, such as the oxygenation of the patient&#39;s blood. 
         [0007]    The ability of any PPG system to accurately and precisely measure the parameter of interest is based on the signal-to-noise ratio (SNR) of the overall system, considering both its transmitter and receiver. For example, it has been observed that an SNR of at least about 30 dB, for the PPG system as a whole including the transmission channel through patient tissue, is necessary in order to measure pulse rate to an accuracy of 1 beat per minute (bpm). A complicating factor in practice is that the system SNR depends on the perfusion index of the patient, as illustrated in  FIG. 2 . Perfusion index is the ratio of the AC signal due to pulsatile blood flow to the DC background level of the light signal passing through the patient&#39;s peripheral tissue, and depends largely on the health and physical condition of the patient. As such, for the PPG system of  FIG. 1 , the perfusion index is reflected in the amplitude of the AC pulses of the received light (e.g., as output by amplifier  8 ) relative to its DC level. As shown in  FIG. 2 , the system SNR increases at higher perfusion index values. 
         [0008]    Conversely, if the SNR of the PPG system can be increased, the system can measure the pulse rate and blood oxygenation in a wider range of patients, particularly those of poorer health and thus lower perfusion indices.  FIG. 2  shows two SNR vs. perfusion index plots  9   a ,  9   b . Plot  9   a  illustrates the relationship of system (transmit—channel—receive) SNR to perfusion index for the case in which the transmitter SNR is 95 dB; as known in the art, to attain the required system SNR, the SNRs for the transmitter and receiver must both be higher than that required system SNR. At that transmit SNR, the PPG system is able to measure pulse rate to an accuracy of one bpm only for patients exhibiting a perfusion index of at least about 0.06. In contrast, plot  9   b  illustrates that if the transmitter is able to operate at an SNR of 110 dB, pulse rate measurements at an error of 1 bpm can be made for patients with perfusion index values as low as about 0.01. Accordingly, noise in the transmitter of the PPG system is a critical factor in covering a wide range of patients. 
         [0009]    So-called “wearable” electronic devices, such as fitness monitoring devices, have recently become popular. In addition to fitness monitoring devices, wearable medical monitoring devices are being contemplated for use in healthcare, for example to monitor the recovery or progress of a patient suffering from a medical condition. As such, the use of PPG to obtain oxygenation, pulse rate, and other measurements by way of a wearable device, particularly such a device that can be worn all day, is desirable. In this context, battery life becomes of critical performance. 
         [0010]    In this regard, an important electrical parameter of a transmitter in a battery-powered system, such as a wearable device, is the “headroom” of the LED driver. As well-known by those in the art, it is desirable that battery-powered systems operate at low power supply voltages to reduce power consumption and to reduce the cost of the battery itself. In transmitter  3  of  FIG. 1 , the voltage drop across LED  2  in its operating state is defined by its material. The headroom, shown as V head  in  FIG. 1 , is the voltage required by LED driver  4  beyond the LED voltage drop. Conversely, the minimum V dd  power supply voltage is the sum of the voltage drop across LED  2  and the headroom V head  of LED driver  4 . For battery powered systems, therefore, it is desirable that the headroom V head  required by LED driver  4  be minimized, especially considering that the output voltage from conventional batteries tends to sag over time. 
         [0011]    As mentioned above, it is desirable to minimize power consumption in PPG systems, particularly those in battery-powered wearable devices intended for “all-day” use. Duty cycling of LED driver  4  in transmitter  3  is a common approach to reducing system power consumption. It is therefore desirable for LED driver  4  to exhibit fast switching, and rapid settling times, so that the “on” pulse width can be reduced as much as possible and thus minimizing power consumption. 
         [0012]      FIGS. 3 a  through 3 c    illustrate examples of conventional LED drivers for PPG systems. The circuit of  FIG. 3 a    is an example of a typical LED driver circuit, such as used in a PPG system as described above. In this circuit, LED  10  has its anode at the V dd  power supply voltage and its cathode connected to the drain of rise time control n-channel MOS transistor  12 . Transistor  12  has its source connected to the drain of n-channel driver transistor  14 , which is connected to ground via variable resistor  16 . Resistor  16  operates to control the current drawn through LED  10  and transistor  12 , at a resistance typically set by a digital-to-analog converter (DAC). The gate of transistor  12  receives a control voltage from rise time controller  13 , which is an adjustable circuit block that controls the conduction of transistor  12  to attain the desired rise and fall times in the turn-on and turn-off characteristics of LED  10 . Amplifier  18  receives a reference voltage VREF at its non-inverting (positive) input and a feedback voltage from the source of transistor  14  at its inverting (negative) input. Output voltage VGATE from amplifier  18  is applied to the gate of driver transistor  14 . According to this arrangement, amplifier  18  operates to drive the gate voltage VGATE at driver transistor  14  so that reference voltage VREF at the source node of transistor  14 . Reference voltage VREF is modulated to selectively forward bias LED  10 . 
         [0013]    The LED driver circuit of  FIG. 3 a    provides certain advantages in a PPG system. Specifically, variable resistor  16  tends to reduce the transmitter noise in this circuit, and the ripple exhibited by this circuit is also quite low. However, it has been observed that this arrangement is vulnerable to significant input flicker noise from amplifier  16 , degrading transmitter performance. The settling time of this LED driver is also quite slow, due to the bandwidth of amplifier  18 . In addition, the LED driver of  FIG. 3 a    is not conducive to implementation in low voltage, battery-powered, applications because of its large headroom voltage, specifically the sum of the drain-to-source overdrive voltages of transistors  12  and  14  plus the voltage drop across resistor  16 . 
         [0014]      FIG. 3 b    illustrates a conventional LED driver with very low headroom requirements as useful for a PPG system. In this circuit, power supply  20  applies the V dd  bias to LED  10  through inductor  22 . N-channel driver transistor  26  has its source-drain path connected in parallel with LED  10  between inductor  22  and ground. The gate of transistor  26  receives the output of pulse-width modulator (PWM)  24 . During the “off” pulses, transistor  26  shunts the inductor current through inductor  22  to ground; during the “on” pulses, the V dd  power supply voltage forward biases LED  10 , such that the inductor current is conducted through LED  10  to ground.  FIG. 3 b    illustrates the behavior of output current I out  through LED  10  over a sequence of pulses from PWM  24 , illustrating that the output current I out  appears as a sequence of triangle waves. While the headroom required by this LED driver is quite low, it has been observed that transmitter noise is quite high in this arrangement, which reduces the patient coverage as discussed above relative to  FIG. 2 . In addition, significant ripple is present in the LED driver of  FIG. 3   b.    
         [0015]    In the circuit of  FIG. 3 c   , LED  10  has its anode at the V dd  power supply voltage and its cathode connected to the drain of n-channel MOS transistor  14 ; the source of transistor  14  is at ground. A reference voltage VREF is applied to the gate of transistor  14 , and is modulated to turn LED  10  on and off, thus controlling the emission of light. This simple driver of  FIG. 3 c    has a low headroom voltage of only the drain-to-source voltage overdrive of transistor  14 , and exhibits no ripple. However, because the driver of  FIG. 3 c    is quite noisy, its use in the transmitter of a PPG system will have limited patient coverage, as discussed above relative to  FIG. 2 . 
         [0016]    By way of further background, auto-zeroing techniques for removing offset voltage and drift of operational amplifiers are known in the art, as described in Kugelstadt, “Auto-zero amplifiers ease the design of high-precision circuits”,  Analog Applications Journal,  2Q 2005 (Texas Instruments Incorporated), pp. 19-28, incorporated herein by reference. 
       BRIEF SUMMARY OF THE INVENTION 
       [0017]    Disclosed embodiments provide an LED driver that reduces transmitter noise, when incorporated into a photoplethysmography (PPG) system. 
         [0018]    Disclosed embodiments provide such an LED driver that reduces the transmitter settling time, by reducing LED turn-on time, and is thus capable of operating with lower power consumption than conventional LED driver circuits. 
         [0019]    Disclosed embodiments provide such an LED driver that has minimal headroom voltage requirements. 
         [0020]    Disclosed embodiments provide such an LED driver that, when implemented in the transmitter of a pulse oximeter or heart rate monitor, increases the patient coverage with respect to perfusion index. 
         [0021]    Other objects and advantages of the disclosed embodiments will be apparent to those of ordinary skill in the art having reference to the following specification together with its drawings. 
         [0022]    According to certain embodiments, an LED driver circuit, such as useful for use in a photoplethysmography (PPG) system, includes a driver transistor connected in series with a variable current control resistor and the LED itself. A switched-mode amplifier circuit drives the gate of the driver transistor in two non-overlapping clock phases during an “on” pulse. In a first phase, noise and offset of the amplifier is stored as a voltage by a capacitor coupled between an amplifier input and the output of the amplifier. Also in this first phase, a precharge capacitor between the gate of the driver transistor and a ground node is precharged to a reference voltage level. In a second phase, the voltage applied to the amplifier input is compensated by the stored sampled noise voltage and is not reflected at the gate voltage of the driver transistor as it turns on the LED. The voltage swing of the driver gate voltage in this second phase is reduced by the precharge applied in the first phase. 
         [0023]    According to another embodiment, a PPG system is constructed to include a transmitter with an LED driver that is co-located with a photodiode receiver. A linear estimate of ripple and noise current in the transmitter is forwarded to a noise and ripple remover circuit in the receiver to recover the signal component of the light signal received by the photodiode. 
     
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWING 
         [0024]      FIG. 1  is an electrical diagram, in schematic form, of a generic architecture for a conventional photoplethysmography (PPG) system. 
           [0025]      FIG. 2  illustrates plots of signal-to-noise ratio (SNR) of conventional PPG systems having different transmitter SNRs versus perfusion index. 
           [0026]      FIGS. 3 a  through 3 c    are electrical diagrams, in schematic form, of conventional LED drivers for PPG systems. 
           [0027]      FIG. 4  is an electrical diagram, in block form, of a PPG system in which embodiments may be implemented. 
           [0028]      FIG. 5  is an electrical diagram, in schematic form, of a transmitter in a PPG system including an LED driver constructed according to an embodiment. 
           [0029]      FIG. 6 a    is a timing diagram illustrating clock phases in the operation of the LED driver of  FIG. 5  according to an embodiment. 
           [0030]      FIGS. 6 b  and 6 c    are electrical diagrams, in schematic form, illustrating the operation of the LED driver of  FIG. 5  in respective clock phases as shown in  FIG. 6   a.    
           [0031]      FIG. 7 a    is an electrical diagram, in block and schematic form, of a PPG system constructed according to another embodiment. 
           [0032]      FIG. 7 b    is a plot of forward current versus voltage for the LED driven by an LED driver according to disclosed embodiments. 
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0033]    The one or more embodiments described in this specification are implemented into a photoplethysmography (PPG) system such as used in pulse oximetry, as it is contemplated that such implementation is particularly advantageous in that context. However, it is also contemplated that concepts of this invention may be beneficially applied to other applications involving the transmission and receipt of optical signals. Accordingly, it is to be understood that the following description is provided by way of example only, and is not intended to limit the true scope of this invention as claimed. 
         [0034]    Referring now to  FIG. 4 , the construction and operation of a PPG system, for example serving as a pulse oximeter and heart rate monitor, and in which these embodiments may be implemented will be described. As shown in  FIG. 4 , this system includes analog front end (AFE)  30 , in which both a transmitter channel and a receiver channel are implemented. The transmitter channel of AFE  30  operates to drive one or more light-emitting diodes (LEDs)  40  connected externally to AFE  30 . For the example of a pulse oximeter, LEDs  40  are constituted as a pair of LEDs of different colors, for example red and infrared, or red and green. An example of a conventional AFE for pulse oximeters and heart rate monitor is the AFE 4400 integrated AFE available from Texas Instruments Incorporated, described in “AFE4400 Integrated Analog Front-End for Heart Rate Monitors and Low-Cost Pulse Oximeters”, Data Sheet SBAS601H (Texas Instruments Incorporated, 2014), incorporated herein by this reference. 
         [0035]    The receiver channel of AFE  42  operates to detect and process electrical signals from photodiode receptor  42 , which is also connected externally to AFE  30 . In this example, receiver channel of AFE  30  according to this embodiment includes amplifier  44 , which is connected to external photodiode receptor  42  and which operates to amplify the signal provided by receptor  42 . The output of amplifier  44  is coupled to the input of analog-to-digital converter (ADC)  46  (via optional noise remover  45 , which will be further described below according to an embodiment), which processes and digitizes the amplified photodiode signal in the conventional manner. Interface  31  communicates that received signal to the appropriate processor circuitry (not shown) for analysis and display of the various attributes indicated by the received signal. 
         [0036]    The transmitter channel of AFE  30  in this example includes LED current control digital-to-analog converter (DAC)  34 , which receives signals indicative of the desired operating mode and functionality of LEDs  40 , from external processor or other circuitry via interface  31 . In response, LED current control DAC  34  forwards control signals to LED driver  36 , which in turn applies the desired excitation to LEDs  40 , specifically by controlling the forward bias current of LEDs  40  and correspondingly the intensity of the emitted light. LED driver  36  may also optionally provide signal to an optional noise/ripple remover  45  in the receiver channel, as will be further described below in connection with an additional embodiment. 
         [0037]    AFE  30  also includes timing controller  32 , which at least in part controls the operation of transmitter and receiver channels. Other power, diagnostic, and control functions may also be provided within AFE  30 , as conventional in the art. 
         [0038]    Referring now to  FIG. 5 , the construction of LED driver  36  according to an embodiment will now be described. In this embodiment, LED driver  36  controls the light emitted by LED  40  by applying a voltage VGATE at the gate of n-channel metal-oxide-semiconductor (MOS) driver transistor  54 . While a single LED  40  is illustrated in the schematic diagram of this  FIG. 5 , it is contemplated that this LED  40  may be one of a pair of LEDs such as used in a pulse oximeter; in that case, the schematic of  FIG. 5  illustrates the manner in which that one of the pair of LEDs is driven in operation. It is contemplated that those skilled in the art having reference to this specification will be readily arrange LED driver  36  to drive the other LED in the pair, for example according to one of the arrangements described in the above-incorporated AFE4400 data sheet. 
         [0039]    In the arrangement of  FIG. 5 , the anode of LED  40  is biased at the V dd  power supply voltage, and its cathode is coupled by switch  52  to the drain of driver transistor  54 . The source of transistor  54  is coupled to ground via variable current control resistor  56 . The resistance of this current control resistor  56  is controlled by LED current control DAC  34 , which as mentioned above operates to control the amplitude of the light emitted by LED  40 . In this embodiment as shown in  FIG. 5 , capacitor  70  is present between the gate of driver transistor  54  and the ground node. It is contemplated that this capacitor  70  will typically be constituted by the parasitic gate-to-body node capacitance of driver transistor  54 , but if desired or necessary for the functionality described below, may be supplemented by an additional capacitor. 
         [0040]    LED driver  36  receives reference voltage VREF, as may be applied from externally to AFE  30  or generated within AFE  30  by a conventional bandgap circuit or other voltage reference circuit. As will be evident from the following description, reference voltage VREF determines the voltage VGATE applied to the gate of driver transistor in its “on” state, that is when LED  40  is to be emitting light. The input receiving reference voltage VREF is connected through switch  61  to one plate of capacitor  60 , at sample node SMP, and through switch  65  to an opposite plate of capacitor  60  and the non-inverting (positive) input of amplifier  58 . Capacitor  62  is connected between the VREF input and ground, to absorb noise. The output of amplifier  58  is connected to sample node SMP through switch  63 , to the inverting (negative) input of amplifier  58  through switch  67 , and to the gate of n-channel driver transistor  54 . Switch  69  couples the source of driver transistor  54  to the inverting input of amplifier  58 . 
         [0041]    Each of switches  52 ,  61 ,  63 ,  65 ,  67 ,  69  of LED driver  36  is controlled by one of two clock phases Φ1 and Φ2 generated by timing controller  32  ( FIG. 4 ). Each of switches  52 ,  61 ,  63 ,  65 ,  67 ,  69  may be realized as a MOS pass gate, for example realized as a single MOS transistor receiving the corresponding gate voltage corresponding to its clock phase, or as paired p-channel and n-channel MOS transistors with their source-drain paths connected in parallel and receiving complementary gate voltages. 
         [0042]    Specifically, switches  63 ,  65 , and  67  are controlled by clock phase Φ1, specifically to be closed when clock phase Φ1 is at an active high level, while switches  52 ,  61 , and  69  are similarly controlled by clock phase Φ2.  FIG. 6 a    illustrates the relationship of clock phases Φ1 and Φ2 as generated by timing controller  32  in this embodiment. As shown in  FIG. 6 a   , these clock phases Φ1 and Φ2 are non-overlapping clock pulses that are driven to an active high level during the duration of an enable signal applied to amplifier  58 , which corresponds to the duration of light emission from LED  40 . 
         [0043]    According to this embodiment, LED driver  36  operates in clock phase Φ1 to precharge the gate voltage VGATE of driver transistor  54  to a voltage about at the level of reference voltage VREF, and to sample the offset and “flicker” noise of amplifier  58 ; LED  40  is not driven to emit light during clock phase Φ1. LED driver  36  operates in clock phase Φ2 to drive amplifier  58  with an input voltage corresponding to reference voltage VREF, compensated for the offset and noise sampled during clock phase Φ1. Driver transistor  54  is turned on by the voltage VGATE at the output of amplifier  58  in response to this compensated voltage at its non-inverting input, and LED  40  emits light accordingly. As will be evident from this description, this noise compensation at the voltage applied to amplifier  58  minimizes the undesired noise and ripple that is typically present in conventional LED driver circuits with low headroom requirements. 
         [0044]      FIGS. 6 b  and 6 c    illustrate the operation of LED driver  36  in clock phases Φ1 and Φ2, respectively. In  FIG. 6 b   , switches  63 ,  65 , and  67  are closed during clock phase Φ1, and switches  52 ,  61 , and  69  are open. As such, the VREF input is connected to the non-inverting input of amplifier  58  via switch  65 , and sample node SMP at the opposing plate of capacitor  60  is connected to the output of amplifier  58  via switch  63 . The output of amplifier  58  is also directly connected to the inverting input of amplifier  58  via switch  67 . The cathode of LED  40  is disconnected from the drain of driver transistor  54 , and as such LED  40  emits no light during clock phase Φ1. 
         [0045]    With LED driver  36  in the state shown in  FIG. 6 b    during clock phase Φ1, amplifier  58  is in unity gain mode, since its output is connected directly to its inverting input through closed switch  67 . This unity gain arrangement causes amplifier  58  to drive its output to reference voltage VREF at its non-inverting input. Sampling capacitor  60 , connected between the non-inverting input and the output of amplifier  58 , will charge to a voltage V SMP  corresponding to the offset plus “flicker” or 1/f noise of amplifier  58 . Meanwhile, capacitor  70  charges to the voltage at the output of amplifier  58 , which as noted above, is at reference voltage VREF, which in turn precharges the gate of driver transistor  54  to about the level of VREF, in anticipation of clock phase Φ2. 
         [0046]    In clock phase Φ2, LED driver  36  is in the state shown in  FIG. 6 c   , with switches  63 ,  65 , and  67  open and switches  52 ,  61 , and  69  closed. The cathode of LED  40  is connected to the drain of driver transistor  54  through the closed switch  52 , such that the bias condition of LED  40 , and thus whether it emits light, is controlled by the state of driver transistor  54 . The inverting input of amplifier  58  is now connected to the source of driver transistor  54 , at the node between transistor  54  and variable resistor  56 , since switch  69  is closed and switch  67  is open in this clock phase Φ2. And the VREF input is coupled to the non-inverting input of amplifier  58  via capacitor  60 , by the action of switch  61  being closed and switch  65  being open. 
         [0047]    According to this embodiment, therefore, the voltage applied to the non-inverting input of amplifier  58  is reference voltage VREF minus the voltage V SMP  stored across capacitor  60  during clock phase Φ1. Since this voltage V SMP  across capacitor  60  corresponds to the offset and noise of amplifier  58 , as sampled in clock phase Φ1, the input voltage applied to the non-inverting input of amplifier  58  is compensated for this offset and noise. The drive applied by amplifier  58  as gate voltage VGATE to driver transistor  54  is thus compensated for offset and noise, which in turn compensates the intensity of the light emitted by LED  40  for the offset and noise at amplifier  58 . By reducing the noise in the emitted light in this fashion, the transmitter SNR is improved according to this embodiment. 
         [0048]    Because the “flicker” noise of typical amplifiers, such as an op amp or the like used to realize amplifier  58 , is primarily low frequency noise (i.e., flicker noise often being referred to as 1/f noise) and because offset is essentially at DC, little variation is expected between the offset plus noise when sampled during clock phase Φ1, and when compensated during clock phase Φ2. It is therefore contemplated that the accuracy of this compensation will generally be quite good. The noise and offset performance of LED driver  36  according to this embodiment is therefore contemplated to be significantly improved over conventional LED driver circuits, particularly those with low headroom as described above. 
         [0049]    In addition, the precharging of gate voltage VGATE to the reference voltage VREF during clock phase Φ1, according to this embodiment, is contemplated to significantly reduce the voltage swing required at the gate of driver transistor  54  when turning on LED  40  in clock phase Φ2. It is expected that the voltage at the source of driver transistor  54  will generally be near reference voltage VREF, on one side or the other, depending on the desired current as controlled via variable resistor  56 . Accordingly, the reduced voltage swing of gate voltage VGATE provided by this precharging is expected to greatly reduce the settling time of LED driver  36 , enabling operation of the PPG system at shorter “on” pulse widths, and thus saving battery power. 
         [0050]    In addition, the headroom requirement of LED driver  36  according to this embodiment is reduced from conventional arrangements such as that described above relative to  FIG. 3 a   . Referring to  FIG. 5 , the headroom required in LED driver  36  is the sum of the voltage drop across variable resistor  56  plus the drain-to-source overdrive of driver transistor  54 . Conversely, the conventional LED driver of  FIG. 3 a    requires headroom that includes the overdrive of two MOS transistors. Accordingly, this embodiment provides an LED driver having relatively low headroom requirements as compared with conventional drivers, while also achieving good noise performance and also fast settling times. 
         [0051]    According to another embodiment, a PPG system is provided that is constructed and operated to provide additional compensation for transmitter noise and ripple. It is contemplated that this embodiment will be especially beneficial when implemented in PPG systems in which transmitter  80  and receiver  85  are “co-located”, for example integrated into the same integrated circuit as one another, or at least in very close proximity with one another. 
         [0052]      FIG. 4  illustrates this embodiment in a general sense by way of optional noise/ripple remover function  45  in the receiver channel of AFE  30 , between amplifier  44  and ADC  46 . As shown in  FIG. 4 , noise/ripple remover  45  receives time-domain signal I_EST from LED driver  36 . This signal I_EST corresponds to the current conducted by LED  40  in its on state, including ripple and other transmitter noise in that current. Because the intensity of the light emitted by an LED is proportional to the forward-bias current, it is contemplated, according to this embodiment, that variations due to noise and ripple in the current applied by LED driver  36  will reflect transmitter noise in the light emitted by LED  40 . 
         [0053]      FIGS. 7 a  and 7 b    illustrate an example of a PPG system including transmitter channel  80  and receiver channel  85  constructed according to this embodiment. Transmitter channel  80  in this PPG system includes LED  82  and LED driver  84  connected in series between the V dd  power supply voltage and ground. LED driver  84  may be constructed in the manner described above relative to  FIGS. 5 and 6   a  through  6   c ; alternatively, LED driver  84  may be constructed according to one of any other conventional LED driver arrangements. In this embodiment, signal I_EST corresponding to the current conducted by LED  82  is communicated from transmitter  80  to receiver  85 , in the form of a sensed voltage at a node in the LED current path from which the current can be deduced, or as a signal corresponding to a direct measure of the conducted current. For example, if LED driver  84  is constructed as described above relative to  FIG. 5 , this sensed voltage may be taken across resistor  56 , or across a small linear current sensing resistor (not shown) in the LED current path. 
         [0054]    Receiver channel  85  includes photodiode  86  with its cathode at the V dd  power supply and its anode connected to an input of amplifier  88 , as conventional in the art. According to this embodiment, however, and as also illustrated in  FIG. 4  discussed above, receiver channel  85  includes noise/ripple remover function  90  (which corresponds to noise/ripple remover function  45  of  FIG. 4 ). Noise/ripple remover  90  may be constructed by way of the appropriate analog circuits for carrying out the functions described in this specification. In the embodiment of  FIG. 4 , noise/ripple remover  90  has one input coupled to the output of amplifier  88  to receive the amplified received signal from photodiode  86 , and another input receiving the voltage or current signal from transmitter channel  80 . As will be described in further detail below, noise/ripple remover  90  removes the estimated transmitter noise in the LED current, and thus in the emitted light, from the amplified signal received from photodiode  86 . The output of noise/ripple remover  90  is then forwarded along receiver channel  85  for further processing and communication. Alternatively, noise/ripple remover  45  may be realized in the digital domain (i.e., following ADC  46 ) as a digital circuit or programmable logic, arranged or programmed to carry out these functions and operations in the digital domain. Further in the alternative, some of the functions and operations of noise/ripple remover  90  may be performed in the analog domain and others in the digital domain following digitization by ADC  46 . In any event, it is contemplated that those skilled in the art having reference to this specification will be readily able to implement noise/ripple remover  90  as appropriate to carry out these functions in particular applications, in either the analog or digital domains or in a combination of the two, without undue experimentation. 
         [0055]    The operation of noise/ripple remover  90  according to this embodiment will now be described in connection with the manner in which the noise and ripple removal function is performed. One may consider the current I TX  conducted by LED  82  as the sum of a signal component I TX0  and a noise component I Tx,n : 
         [0000]    
       
      
       I 
       TX 
       =I 
       TX0 
       +I 
       TX,n  
      
     
         [0000]      FIG. 7 b    is a current-voltage characteristic of a typical LED that may be used as LED  82  in transmitter  80 . As shown in this Figure, at applied voltages above some threshold voltage V t , forward current I fwd  conducted by the LED is linear with the applied voltage V, to at least a reasonable approximation. Accordingly, it is contemplated that a voltage V d  taken at the cathode of LED  82  in transmitter  80  of  FIG. 7 a   , and communicated to noise/ripple remover  90  as signal I_EST, will similarly be linear with the voltage drop across LED  82 , at a slope corresponding to a small-signal (“AC”) resistance R d . Accordingly, the voltage V d  can be considered as the sum of a nominal signal voltage component V d0  and a noise component: 
         [0000]    
       
      
       V 
       d 
       =V 
       d0 
       +R 
       d 
       I 
       TX,n  
      
     
         [0000]    According to this embodiment, noise/ripple remover function  90  determines nominal signal voltage component V d0  by time-averaging the voltage V d  communicated by transmitter channel  80  as signal I_EST. The AC resistance R d  may be determined by characterization or the specifications of LED  82 , and programmed or otherwise set at noise/ripple remover function  90  for use in this determination. 
         [0056]    The light emitted by LED  82  will be proportional to the current I TX  conducted by LED  82 , including both the signal and noise components noted above. At receiver channel  85 , the voltage V RX  at the output of amplifier  88  corresponding to the signal from photodiode  86  can thus be expressed as: 
         [0000]    
       
      
       V 
       RX 
       ={tilde over (G)}I 
       TX  
      
     
         [0000]    where {tilde over (G)} is the transfer function of the medium through which the emitted light passes between LED  82  and photodiode  86  (e.g., the patient&#39;s finger in a pulse oximeter application). Breaking down the LED current I TX  into its signal and noise components: 
         [0000]    
       
      
       V 
       RX 
       ={tilde over (G)}I 
       TX0 
       +{tilde over (G)}I 
       TX,n  
      
     
         [0000]    which can also be expressed in the voltage domain as the sum of signal and noise components: 
         [0000]    
       
      
       V 
       RX 
       =V 
       RX0 
       +V 
       RX,n  
      
     
         [0000]    Signal component V RX0  is the desired quantity to be processed: 
         [0000]    
       
         
           
             
               V 
               
                 RX 
                  
                 
                     
                 
                  
                 0 
               
             
             = 
             
               
                 
                   V 
                   RX 
                 
                 - 
                 
                   V 
                   
                     RX 
                     , 
                     n 
                   
                 
               
               = 
               
                 
                   V 
                   RX 
                 
                  
                 
                   ( 
                   
                     1 
                     - 
                     
                       
                         V 
                         
                           RX 
                           , 
                           n 
                         
                       
                       
                         V 
                         RX 
                       
                     
                   
                   ) 
                 
               
             
           
         
       
     
         [0000]    The ratio of noise V RX,n  to received voltage V RX  can be expressed as: 
         [0000]    
       
         
           
             
               
                 V 
                 
                   RX 
                   , 
                   n 
                 
               
               
                 V 
                 RX 
               
             
             = 
             
               
                 
                   
                     G 
                     ~ 
                   
                    
                   
                     I 
                     
                       TX 
                       , 
                       n 
                     
                   
                 
                 
                   
                     G 
                     ~ 
                   
                    
                   
                     I 
                     TX 
                   
                 
               
               = 
               
                 
                   I 
                   
                     TX 
                     , 
                     n 
                   
                 
                 
                   I 
                   TX 
                 
               
             
           
         
       
     
         [0000]    Since the transmitted current noise component I TX,n  is: 
         [0000]    
       
         
           
             
               I 
               
                 TX 
                 , 
                 n 
               
             
             = 
             
               
                 
                   V 
                   d 
                 
                 - 
                 
                   V 
                   
                     d 
                      
                     
                         
                     
                      
                     0 
                   
                 
               
               
                 R 
                 d 
               
             
           
         
       
     
         [0000]    the signal component V RX0  can be determined as: 
         [0000]    
       
         
           
             
               V 
               
                 RX 
                  
                 
                     
                 
                  
                 0 
               
             
             = 
             
               
                 V 
                 RX 
               
                
               
                 ( 
                 
                   1 
                   - 
                   
                     
                       
                         V 
                         d 
                       
                       - 
                       
                         V 
                         
                           d 
                            
                           
                               
                           
                            
                           0 
                         
                       
                     
                     
                       
                         R 
                         d 
                       
                        
                       
                         I 
                         TX 
                       
                     
                   
                 
                 ) 
               
             
           
         
       
     
         [0000]    Because the received signal V RX  from photodiode  86 , the signal I_EST from transmitter  80  in the form of voltage V d , a time-average of that voltage V d  as an estimate of nominal voltage V d0 , the AC resistance R d , and the current I TX  as set by LED current control DAC  32  are all available to noise/ripple remover function  90 , this equation can be solved by function  90  to determine the voltage signal component V RX0 . In one implementation example, the subtraction of V d −V d0  may be performed in the analog domain by fixed amplification, followed by digitization (ADC  46 ) of the difference and completion of the calculation of V RX0  in the digital domain. Signal component V RX0 , from which the effects of noise and ripple are removed according to the foregoing calculations, is then forwarded along receiver channel  85  for processing as desired for the particular application of the PPG system. 
         [0057]    Alternatively, as shown in  FIG. 7 a    and as mentioned above, a small resistor  89  may be inserted in series with LED  82  and LED driver  84  to provide a direct measure of current conducted by LED  82  to noise/ripple remover function  90  at receiver  85 . In this approach, signal I_EST will correspond to a measured voltage across resistor  89 . Further in the alternative, a non-contact current sensor (not shown) may be deployed at transmitter  80  to sense the current conducted during the “on” time of LED  82  and communicate signal I_EST corresponding to that measured current to noise/ripple remover  90  in receiver  85  as shown. According to these alternative current measurements, the estimate of the AC resistance R d  of LED  82  is not necessary, allowing the determination of the signal component of the photodiode signal V RX0  in a more direct fashion, without the potential inaccuracy from estimating this resistance. 
         [0058]    It is therefore contemplated that this embodiment can further reduce the effect of transmitter noise in the eventual output signal from the receiver channel of a PPG system. This noise reduction is reflected in an improvement of the overall SNR of the PPG system itself. As discussed above relative to  FIG. 2 , improvement of the system SNR by improving the effective SNR of the transmitter channel (in this embodiment, by compensating for transmitter noise), expands the coverage of the system to patients with lower perfusion index values, and thus expanding the potential applications of PPG in patient monitoring and other applications. 
         [0059]    While one or more embodiments have been described in this specification, it is of course contemplated that modifications of, and alternatives to, these embodiments, such modifications and alternatives capable of obtaining one or more the advantages and benefits of this invention, will be apparent to those of ordinary skill in the art having reference to this specification and its drawings. It is contemplated that such modifications and alternatives are within the scope of this invention as subsequently claimed herein.