Abstract:
In accordance with the present invention, there is provided a method of generating, for a given natural q, a set of q complex sequences of a constant envelope DFT-transformed into sequences also of a constant envelope. It is shown that these sequences are transformed into themselves upon applying the DFT twice. A digital communications method with enhanced bandwidth and transmitted power utilization employing these sequences as basic transmission objects is described. Finally, a method of generating complex sequences identical to their discrete spectra is described.

Description:
FIELD OF THE INVENTION  
       [0001]     The present invention relates to designing new waveforms with special properties for digital communications and, more particularly, to developing complex waveforms of a constant envelope having spectra of a constant envelope and complex waveforms shaped like their spectra.  
       BACKGROUND OF THE INVENTION  
       [0002]     One of the biggest challenges of modern digital communications is designing signal waveforms with improved transmission properties. A properly chosen waveform can substantially enhance the bandwidth utilization, and/or reduce the needed transmission power, and/or improve the peak-to-average power ratio in a communications system.  
         [0003]     Recently, we identified and described a new class of waveforms termed “flat spectrum chirps” (FSC) (Mitlin, 2004). They are represented by complex sequences of a constant envelope mapped by the discrete Fourier transform (DFT) onto complex sequences also of a constant envelope. It was mathematically proven in (Mitlin, 2004) that they are optimal phase shifters capable of greatly reducing peak-to-average power ratios in multicarrier communications systems.  
         [0004]     Flat spectrum chirps are perfect spreading sequences evenly occupying the bandwidth slot where the transmission occurs so that for a given bandwidth the transmission power would be minimized by using FSC. Therefore, they appear to have much promise if served as basic transmission units in a digital communications system.  
         [0005]     This patent application further develops the concept of FSC. It explores another amazing property which we have just discovered, e.g. an FSC transforms into itself upon applying the DFT twice. This property allows us, first, to widen the class of flat spectrum waveforms by including discrete spectra of the existing FSCs; and secondly, to introduce a method of generating waveforms that are identical to their discrete spectra. This application describes properties of these waveforms and discusses their possible applications for digital communications.  
         [0006]     The present application is related to the U.S. patent application Ser. No. 10/945,974 titled: “Phase Shifters for Peak-To-Average Power Ratio Reduction in Multi-Carrier Communications Systems”, invented by Vlad Mitlin, filed on 19 Sep. 2004 and owned by the same assignee now and at the time of invention.  
       SUMMARY OF THE INVENTION  
       [0007]     In accordance with the present invention, there is provided a method of generating, for a given natural q, a set of q complex sequences of a constant envelope DFT-transformed into sequences also of a constant envelope. It is shown that these sequences are transformed into themselves upon applying the DFT twice. A digital communications method with enhanced bandwidth and transmitted power utilization employing these sequences as basic transmission objects is described. Finally, a method of generating of complex sequences identical to their discrete spectra is described. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0008]     A complete understanding of the present invention may be obtained by reference to the accompanying drawings, when considered in conjunction with the subsequent, detailed description, in which:  
         [0009]      FIG. 1  is a dependence R(D) where R is given by Eq. (16) and D is given by Eq. (17);  
         [0010]      FIG. 2  shows the real part of an FSC at q=4 and p=1;  
         [0011]      FIG. 3  shows the imaginary part of an FSC at q=4 and p=1;  
         [0012]      FIG. 4  shows the real part of an FSC at q=4 and p=3;  
         [0013]      FIG. 5  shows the imaginary part of an FSC at q=4 and p=3;  
         [0014]      FIG. 6  shows the real part of a dual FSC at q=4 and p=1;  
         [0015]      FIG. 7  shows the imaginary part of a dual FSC at q=4 and p=1;  
         [0016]      FIG. 8  shows the real part of a dual FSC at q=4 and p=3;  
         [0017]      FIG. 9  shows the imaginary part of a dual FSC at q=4 and p=3;  
         [0018]      FIG. 10  is a dependency of the symbol error rate on the SNR in a FSC-based communications system for different sizes of alphabets;  
         [0019]      FIG. 11  shows the absolute values of spectra of an FSC (crosses) and a non-AWGN channel interferer (solid line);  
         [0020]      FIG. 12  presents the real part of a BSSW at q=64 and p=41;  
         [0021]      FIG. 13  presents the imaginary part of a BSSW at q=64 and p=41;  
         [0022]      FIG. 14  shows the real part of a DSSW;  
         [0023]      FIG. 15  shows the real part of another DSSW;  
         [0024]      FIG. 16  presents the distribution of variances of 10,000 randomly chosen normalized DSSW;  
         [0025]      FIG. 17  presents the distribution of PAPR values of 10,000 randomly chosen normalized DSSW;  
         [0026]      FIG. 18  shows the absolute value of a BSSW at q=256 and p=7 at the transmitting station;  
         [0027]      FIG. 19  shows the absolute value of a BSSW at q=256 and p=7 at the receiving station; and  
         [0028]      FIG. 20  shows the absolute value of a BSSW at q=256 and p=7 at the receiving station after performing 10 iterations of Eq. (25). 
     
    
       [0029]     For purposes of clarity and brevity, like elements and components will bear the same designations and numbering throughout the FIGURES.  
       DESCRIPTION OF THE PREFERRED EMBODIMENT  
       [0000]     1. Flat Spectrum Chirps  
         [0030]     In this subsection it will be proven that for a given q, there exists a set z of complex, unit-envelope sequences of the length q having a unit-envelope DFT, i.e.  
                z   -&gt;          =   1     ;            DFT   ⁡     (     z   -&gt;     )            =   1         
     where     
         {     DFT   ⁡     (     z   -&gt;     )       }     =     {       1     q       ⁢       ∑     r   =   0       q   -   1       ⁢       z   r     ⁢     exp   ⁡     (       -   2     ⁢   jπ   ⁢           ⁢     nr   /   q       )             }         
 
         [0031]     This set is defined as follows: 
 
 {right arrow over (z)} ( r )={exp(± jπr   2   p/q )}  (1) 
 
         [0032]     In Eq. (1) the integer r varies between 0 and q−1; integers p and q are mutually prime and have opposite parities; and j 2 =−1. Eq. (1) describes linear chirps that are well known in digital communications. However, a chirp with mutually prime p and q of opposite parities has a unit-envelope DFT as well. To show this let us consider the following sum:  
               S   ⁡     (     p   ,   q     )       =       ∑     r   =   0       q   -   1       ⁢     exp   ⁡     (       -   jπ     ⁢           ⁢     r   2     ⁢     p   /   q       )                 (   2   )             
 
         [0033]     This is termed the Gauss sum and is considered in the number theory.  
         [0034]     First, it will be proven that the absolute value of a Gauss sum, for p and q specified, is independent of p. Some known properties of Gauss sums will be used. The first property is its multiplicaticity: 
 
( q′,q ″)=1 S ( p,q′q ″)= S ( pq′,q ″) S ( pq″,q ′)  (3) 
 
 i.e. at mutually prime q′ and q″ the sum on the l.h.s. of Eq. (3) can be presented as a product of two other sums. Apply the property (3) to the Gauss sum as follows: 
 
 S (1 ,pq )= S ( p,q ) S ( q,p )  (4) 
 
         [0035]     It did not seem to simplify the problem; however, there is another property of Gauss sums called the Schaar&#39;s identity, and it holds for mutually prime p and q of opposite parity. This identity can be presented as follows: 
 
 S *( p,q )=exp( jπ/ 4) √{square root over (q/p)}S ( q,p )  (5) 
 
 where ‘*’ denotes a complex conjugate. Combining Eqs. (4) and (5) yields: 
 
exp( jπ/ 4) √{square root over (q/p)}S (1 ,pq )=| S ( p,q )| 2   (6) 
 
         [0036]     The sum on the l.h.s. of Eq. (6) can be evaluated by using again the Schaar&#39;s identity: 
 
 S (1 ,pq )=exp(− jπ/ 4) √{square root over (pq)}S *( pq, 1)=exp(− jπ/ 4) √{square root over (pq)}   (7) 
 
         [0037]     Introducing Eq. (7) into Eq. (6) yields: 
 
| S ( p,q )| 2   =q   (8) 
 
         [0038]     This is what had to be proven.  
         [0039]     Next, let us show that for p and q of opposite parity, shifting r in this Gauss sum by an integer m does not change it:  
                 ∑     r   =   0       q   -   1       ⁢     exp   ⁡     (       -   jπ     ⁢           ⁢     r   2     ⁢     p   /   q       )         =       ∑     r   =   0       q   -   1       ⁢     exp   ⁡     (       -       jπ   ⁡     (     r   +   m     )       2       ⁢     p   /   q       )                 (   9   )             
 
         [0040]     The property (9) can be easily proven by induction: we just have to show that (9) is true for a unit shift. This can be seen from the following equality:  
                 ∑     r   =   0       q   -   1       ⁢     exp   ⁡     (       -   j     ⁢           ⁢   π   ⁢           ⁢     r   2     ⁢     p   /   q       )         =       ∑     r   =   0       q   -   1       ⁢     exp   ⁡     (       -       jπ   ⁡     (     q   -   1   -   r     )       2       ⁢     p   /   q       )                 (   10   )             
 
 which, for p and q of opposite parity, can be rewritten as follows:  
                 ∑     r   =   0       q   -   1       ⁢     exp   ⁡     (     -     jπ   ⁡     (     qp   -     2   ⁢     p   ⁡     (     1   +   r     )         +         (     1   +   r     )     2     ⁢     p   /   q         )         )         =       ∑     r   =   0       q   -   1       ⁢     exp   ⁡     (       -       jπ   ⁡     (     r   +   1     )       2       ⁢     p   /   q       )                 (   11   )             
 
         [0041]     Now, Eq. (9) is equivalent to the following equation:  
                   ∑     r   =   0       q   -   1       ⁢     exp   ⁡     (         -   2     ⁢   j   ⁢           ⁢   π   ⁢           ⁢     nr   /   q       -     j   ⁢           ⁢   π   ⁢           ⁢     r   2     ⁢     p   /   q         )         =       S   ⁡     (     p   ,   q     )       ⁢     exp   ⁡     (     jπ   ⁢           ⁢     m   2     ⁢     p   /   q       )           ,     
     ⁢     mp   =     n   ⁡     (     mod   ⁢           ⁢   q     )                 (   12   )             
 
         [0042]     Eq. (12) establishes a relation between the DFT of a chirp with parameters p and q specified above and the Gauss sum in Eq. (1). Since p and q are mutually prime, it is always possible, for a given n, to find m such that remainders of dividing mp and n by q are equal. Moreover, for a given n, such m is uniquely found. Therefore, the DFT of such a chirp has a constant envelope of unity:  
                    DFT   ⁡     (     z   -&gt;     )            =              1     q       ⁢       ∑     r   =   0       q   -   1       ⁢     exp   ⁡     (         -   2     ⁢   jπ   ⁢           ⁢     nr   /   q       -     jπ   ⁢           ⁢     r   2     ⁢     p   /   q         )                =              S   ⁡     (     p   ,   q     )              q       =   1               (   13   )             
 
         [0043]     A chirp, whose constant envelope property pertains to the DFT, is termed the flat spectrum chirp (FSC). While the proof was presented for an FSC with a minus sign in Eq. (1), since absolute values of a complex number and its conjugate are equal, the constant envelope property holds for an FSC with a plus sign in Eq. (1) too. Also, as the IDFT matrix is a conjugate to the DFT matrix, the constant envelope property of an FSC pertains to IDFT too.  
         [0000]     2. DFT(DFT(FSC))=FSC  
         [0044]     In this subsection it will be proven that FSCs have another remarkable property, e.g. applying the DFT to an FSC twice yields the same FSC.  
         [0045]     The proof is numerical. Specifically, a Matlab script was written that computes several functions of p and q. The first function is: 
 
 r ( q,p )=1 −sign (( gcd ( q,p )+( q %2)·( p %2)−1) 2 )  (14) 
 
         [0046]     In Eq. (14) gcd stands for the greatest common divisor of two natural numbers; and ‘%’ denotes the computing a remainder of dividing one integer by another. The function r equals 1 for mutually prime p and q of opposite parity, and equals 0 otherwise.  
         [0047]     The second function is: 
 
 d ( q,p )=1 −sign (|{right arrow over (z)}( q,p )− DFT ( DFT ({right arrow over ( z )}( q,p )))| 2 )  (15) 
 
         [0048]     In Eq. (15) the vector z corresponds to a sequence defined by Eq. (1). The function d equals 1 if the double DFT of z coincides with z, and it equals 0 otherwise.  
         [0049]     The third function is:  
               R   ⁡     (     M   ,   Q     )       =       ∑     q   =   M     Q     ⁢       ∑     p   =   1     q     ⁢     r   ⁡     (     q   ,   p     )                   (   16   )             
 
         [0050]     The fourth function is:  
               D   ⁡     (     M   ,   Q     )       =       ∑     q   =   M     Q     ⁢       ∑     p   =   1     q     ⁢     d   ⁡     (     q   ,   p     )                   (   17   )             
 
 In Eqs. (16) and (17) M and Q are positive integers, and M&lt;Q. 
 
         [0051]      FIG. 1  shows the dependence R(D) at M=2 and Q running between 2 and 256 which is the range of most interest for waveform sizes. One can see that R=D for all Q from this interval which constitutes the numerical proof of the statement made in the beginning of this subsection.  
         [0052]     One should note that FSC is not the only chirp transformed to itself upon applying the DFT twice. As an example, at q=6 all chirps given by Eq. (1) have this property, although not all of them correspond to mutually prime p and q of opposite parity. To determine whether a given chirp has this property, a direct computation similar to the one described in this subsection should be performed.  
         [0000]     3. Dual FSC  
         [0053]     It was proven in the first subsection that for an FSC, we have: 
 
| {right arrow over (z)}|= 1 ; |DFT ({right arrow over ( z )})|=1  (18) 
 
         [0054]     It was proven in the second subsection that for an FSC, we have: 
 
 DFT ( DFT ({right arrow over ( z )})= {right arrow over (z)}   (19) 
 
         [0055]     It follows from Eqs. (18) and (19) that if 
 
 {right arrow over (Z)}=DFT ({right arrow over ( z )})  (20) 
 
then 
 
| {right arrow over (Z)}|= 1; |( DFT ({right arrow over ( Z )}))|=1;  (21) 
 
 i.e. if z is an FSC then Z=DFT(z) has the same property that z has, e.g. it is a complex, unit-envelope sequence mapped by the DFT to another complex, unit-envelope sequence. Z is termed the “dual FSC” (DFSC). 
 
         [0056]     Real and imaginary parts of the complete set of FSC and DFSC at q=4 is shown in FIGS.  2  to  9 .  
         [0057]     In a special case of q being a power of 2 and p=1, the real and the imaginary part of the FSC and those of its spectrum are all the same.  
         [0058]     As described in (Mitlin, 2004), complex sequences of constant envelope mapped by DFT to complex sequences of constant envelope are ideal for reducing peak-to-average power ratios of complex waveforms obtained as the q-point IDFT of complex messages of the length q (for example, in orthogonal frequency division multiplexing (OFDM) communications systems). This can be done by multiplying the r-th component of each of these messages by the r-th component of an FSC/DFSC where r is an integer such that 0≦r&lt;q at the transmitter. To retrieve an original message at the receiver the r-th component of the output of the DFT module of the receiver is divided by the r-th component of this FSC/DFSC.  
         [0059]     Furthermore, using a plurality of FSCs/DFSCs can provide a powerful communications security mechanism in the physical layer if messages are consecutively retrieved from a data stream; for each message retrieved, and parameters of the corresponding FSC/DFSC are determined based on the value of a current element of a pseudorandom sequence that is held proprietary by the owner of the data stream.  
         [0000]     4. FSC/DFSC-Based Transmission Method  
         [0060]     If q is a power of 2 there is a set of q/2 FSC corresponding to odd p&lt;q and a set of q/2 DFSC; thus, there are q complex, unit-envelope baseband waveforms having unit-envelope spectra, i.e. uniform spectral densities. This means that for a given bandwidth, these waveforms have minimal possible power (the envelope squared); i.e. they are perfect spreading sequences. They appear to be perfectly suitable to serve as basic transmission units for power and bandwidth efficient data transmission.  
         [0061]     We developed a simulator to prove this concept. We simulated the data transmission over an AWGN baseband channel. Four cases were considered, and in each case 100,000 FSC of the length q=4, 8, 16, and 32 samples, respectively, were transmitted. Accordingly, sets of two, four, eight, and sixteen FSC waveforms were used. FSC sets were mapped to alphabets consisting of two, four, eight, and sixteen symbols, respectively, corresponding to transmission of one, two, three, and four bits per symbol. The detection of a waveform was made by computing Euclidian distances between the received signal and each of FSC from the set used and selecting the symbol corresponding to FSC with the minimum distance.  FIG. 10  shows the symbol error rate versus the SNR for the different sizes of alphabets. Simulation results for a DFSC-based system are similar to those shown in this figure and are not presented here.  
         [0062]      FIG. 11  shows some of the results of another simulation set in which the channel was distorted by a non-AWGN interferer. Its spectrum is shown in  FIG. 11  by a solid line. Data transmission was performed using a set of 32 FSC with q=64. The spectrum of a typical FSC from this set is shown in  FIG. 11  by crosses. 32000 symbols were transmitted at the signal-to-interferer ratio of about −5 dB, and no errors were detected. One can see that the new waveforms have excellent transmission qualities.  
         [0063]     An additional improvement in the detection was attained when the waveform received, z, was pre-processed at the receiver as follows:  
               z   -&gt;     -&gt;         DFT   ⁡     (     DFT   ⁡     (     z   -&gt;     )       )       +     z   -&gt;       2             (   22   )             
 
 prior to computing the Euclidean distances. 
 
 5. Basic Spectrum-Shaped Waveforms 
 
         [0064]     Let us consider another remarkable feature of the FSC/DFSC waveforms. As both an FSC and its corresponding DFSC satisfy Eq. (19), one can write: 
 
 DFT ( {right arrow over (z)}+DFT ({right arrow over ( z )}))= DFT ({right arrow over ( z )})+ DFT ( DFT ({right arrow over ( z )}))= DFT ({right arrow over ( z )})+ {right arrow over (z)}   (23) 
 
 In other words, the sum of an FSC and its corresponding DFSC transforms by the DFT into itself. We termed this sum the “basic spectrum-shaped waveform” (BSSW). 
 
         [0065]     As an example, for q equal to the power of two, there are q/2 different BSSW. The real and imaginary parts of a BSSW at q=64 and p=41 are shown in  FIGS. 12 and 13 .  
         [0000]     6. Derivative Spectrum-Shaped Waveforms  
         [0066]     Here we show that for a given q, one can construct an infinite number of spectrum-shaped waveforms. Specifically, as the DFT is a linear operation, any linear combination of several BSSW will be a spectrum-shaped waveform. We termed them “derivative spectrum-shaped waveforms” (DSSW). These waveforms are defined as follows: 
 
 DSSW=Σa   k   BSSW   k   (24) 
 
         [0067]     In Eq. (24) a k  are arbitrary complex numbers.  
         [0068]     For a given q, there is an innumerous number of different derivative spectrum-shaped waveforms. An example of the real part of the DSSW obtained as a sum of an FSC with q=64 and p=1 and another FSC with q=64 and p=13 is presented in  FIG. 14 . Another example of the real part of a limiting case of DSSW, e.g. an FSC with q=64 and p=1 is shown in  FIG. 15 .  
         [0069]     Statistical properties of DSSW, however, are very similar, for the same q.  FIG. 16  shows the values of variances of 10000 DSSW at q=64, normalized to unit power and generated by randomly choosing the weights a k  such that  
           ∑     k   =   1       q   /   2       ⁢     a   k       =   1       
 
         [0070]      FIG. 17  shows the values of PAPR in this simulation. One can see that variation of each of these parameters over the variety of DSSW is within ten percent.  
         [0071]     One can envision various applications of DSSW; below we will describe just one of them. DSSW can be generated in a large network as individual communication tools for each user. Specifically, DSSW can be generated and distributed among the users to be their authentication waveforms. At the beginning of a communication session between any two users they have to exchange their authentication waveforms to identify themselves. DSSW are suited very well for this purpose because they coincide with their spectra. This allows an enhanced reconstruction of the authentication waveform received, as follows. Upon receiving the waveform s from a peer, a user performs the following transformation:  
               s   →     =         s   →     +     DFT   ⁡     (     s   →     )         2             (   25   )             
 
 at least once.  FIG. 18  shows absolute values of a BSSW at q=256 and p=7. An authentication event was simulated in the AWGN channel with the SNR of 10 dB.  FIG. 19  shows absolute values of the BSSW at the receiver without performing preprocessing using Eq. (25).  FIG. 20  shows absolute values of the BSSW at the receiver after performing 10 iterations of Eq. (25). One can see that using preprocessing routine (25) greatly improves the convergence of the authentication waveform received to the one transmitted. Our numerical experiments show that the convergence does not improve much after about 10 iterations. 
 
         [0072]     Since other modifications and changes varied to fit particular operating requirements and environments will be apparent to those skilled in the art, the invention is not considered limited to the example chosen for purposes of disclosure, and covers all changes and modifications which do not constitute departures from the true spirit and scope of this invention.  
         [0073]     Having thus described the invention, what is desired to be protected by Letters Patent is presented in the subsequently appended claims.