Abstract:
A driving circuit useful in a magnetic inductive coupling wireless communication system is disclosed. The circuit includes an inductor (coil) and capacitor in series selectively coupled to a power source such as a rechargeable battery. The LC circuit is made to resonate in accordance with a Frequency Shift Keying or other protocol. Such resonance produces a voltage across the inductor. This voltage is used to create a first voltage either by tapping into the coil, or by providing a transformer. The first voltage is coupled to the rechargeable battery by a diode. When the circuit resonates, and when the first voltage exceeds the voltage of the power source, the diode turns on, thus shunting excess current back to recharge the rechargeable battery. By use of this circuit, energy is conserved. Additionally, oscillations can be quickly dampened so as to allow the circuit to transmit at high data rates.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This is a continuation application of U.S. patent application Ser. No. 11/780,369, filed Jul. 19, 2007 (now U.S. Pat. No. 9,162,068), which is a continuation-in-part (CIP) application of U.S. patent application Ser. No. 11/778,486, filed Jul. 16, 2007 (abandoned). Priority is claimed to both of these patent applications, and both are incorporated herein by reference in their entireties. 
    
    
     FIELD OF THE INVENTION 
     The present invention relates to a driving circuit useable in a magnetically-coupled telemetry system, and has particular applicability to implantable medical device systems. 
     BACKGROUND 
     Implantable stimulation devices are devices that generate and deliver electrical stimuli to body nerves and tissues for the therapy of various biological disorders, such as pacemakers to treat cardiac arrhythmia, defibrillators to treat cardiac fibrillation, cochlear stimulators to treat deafness, retinal stimulators to treat blindness, muscle stimulators to produce coordinated limb movement, spinal cord stimulators to treat chronic pain, cortical and deep brain stimulators to treat motor and psychological disorders, and other neural stimulators to treat urinary incontinence, sleep apnea, shoulder sublaxation, etc. The present invention may find applicability in all such applications, although the description that follows will generally focus on the use of the invention within a Spinal Cord Stimulation (SCS) system, such as that disclosed in U.S. Pat. No. 6,516,227, which is incorporated herein by reference in its entirety. 
     Spinal cord stimulation is a well-accepted clinical method for reducing pain in certain populations of patients. As shown in  FIGS. 1A and 1B , a SCS system typically includes an Implantable Pulse Generator (IPG)  100 , which includes a biocompatible case  30  formed of titanium for example. The case  30  typically holds the circuitry and power source or battery necessary for the IPG to function, although IPGs can also be powered via external RF energy and without a battery. The IPG  100  is coupled to electrodes  106  via one or more electrode leads (two such leads  102  and  104  are shown), such that the electrodes  106  form an electrode array  110 . The electrodes  106  are carried on a flexible body  108 , which also houses the individual signal wires  112  and  114  coupled to each electrode. In the illustrated embodiment, there are eight electrodes on lead  102 , labeled E 1 -E 8 , and eight electrodes on lead  104 , labeled E 9 -E 16 , although the number of leads and electrodes is application specific and therefore can vary. 
     As shown in  FIG. 2 , the IPG  100  typically includes an electronic substrate assembly  14  including a printed circuit board (PCB)  16 , along with various electronic components  20 , such as microprocessors, integrated circuits, and capacitors mounted to the PCB  16 . Two coils are generally present in the IPG  100 : a telemetry coil  13  used to transmit/receive data to/from an external controller  12  as explained further below; and a charging coil  18  for charging or recharging the IPG&#39;s power source or battery  26  using an external charger (not shown). The telemetry coil  13  can be mounted within the header connector  36  as shown. 
     As just noted, an external controller  12 , such as a hand-held programmer or a clinician&#39;s programmer, is used to send data to and receive data from the IPG  100 . For example, the external controller  12  can send programming data to the IPG  100  to dictate the therapy the IPG  100  will provide to the patient. Also, the external controller  12  can act as a receiver of data from the IPG  100 , such as various data reporting on the IPG&#39;s status. The external controller  12 , like the IPG  100 , also contains a PCB  70  on which electronic components  72  are placed to control operation of the external controller  12 . A user interface  74  similar to that used for a computer, cell phone, or other hand held electronic device, and including touchable buttons and a display for example, allows a patient or clinician to operate the external controller  12 . 
     Wireless data transfer between the IPG  100  and the external controller  12  takes place via inductive coupling, and specifically magnetic inductive coupling. To implement such functionality, both the IPG  100  and the external controller  12  have coils  13  and  17  respectively. Either coil can act as the transmitter or the receiver, thus allowing for two-way communication between the two devices. When data is to be sent from the external controller  12  to the IPG  100  for example, coil  17  is energized with alternating current (AC), which induces an electromagnetic field  29 , which in turn induces a current in the IPG&#39;s telemetry coil  13 . The power used to energize the coil  17  can come from a battery  76 , which like the IPG&#39;s battery  26  is preferably rechargeable, but power may also come from plugging the external controller  12  into a wall outlet plug (not shown), etc. The induced current in coil  13  can then be transformed at the IPG  100  back into the telemetered data signals. To improve the magnetic flux density, and hence the efficiency of the energy transfer, the IPG&#39;s telemetry coil  13  may be wrapped around a ferrite core  13 ′. 
     As is well known, inductive transmission of data from coil  17  to coil  13  can occur transcutaneously, i.e., through the patient&#39;s tissue  25 , making it particular useful in a medical implantable device system. During the transmission of data, the coils  13  and  17  lie in planes that are preferably parallel. Such an orientation between the coils  13  and  17  will generally improve the coupling between them, but deviation from ideal orientations can still result in suitably reliable data transfer. 
     To communicate a serial stream of digital data bits via inductive coupling, some form of modulation is generally employed. In a preferred embodiment, Frequency Shift Keying (FSK) can be employed, in which the logic state of a bit (either a logic ‘0’ or a logic ‘1’) corresponds to the frequency of the induced magnetic field  29  at a given point in time. Typically, this field has a center frequency (e.g., fc=125 kHz), and logic ‘0’ and ‘1’ signals comprise offsets from that center frequency (e.g., f0=121 kHz and f1=129 kHz respectively). Once the data is modulated in this manner at the transmitting device (e.g., the external controller  12 ), it is then demodulated at the receiving device (e.g., the IPG  100 ) to recover the original data. While FSK modulation may be preferred for a given application, one skilled in the art will recognize that other forms of data modulation (e.g., amplitude modulation, On-Off-Keying (OOK), etc.) can be used as well. These modulation schemes as used in a medical implantable device system are disclosed in U.S. Pat. No. 7,177,698, which is incorporated herein by reference in its entirety, and because they are well known, they are not further discussed. 
     A typical driving circuit (or an amplifier circuit)  150  used to energize the transmitting coil is shown in  FIG. 3 . In the example shown it is assumed that the external controller  12  is acting as the transmitter, although it should be remembered that the IPG  100  could also act as the transmitter. The driving circuit  150  comprises a RLC circuit (or as it is sometimes known in the art, a “tank circuit” or “resonant circuit”), in which the inductor (L) comprises the coil  17  in the external controller discussed previously. As one skilled in the art understands, the RLC circuit is made to resonate by the application of a clock signal to switches  152  and  152 ′. Specifically, in the embodiment shown, switches  152  and  152 ′ are toggled out of phase, which is accomplished using clocking signals, Clk and Clk′, which are the inverse or complement of each other. The clock signals are derived from a modulation circuit  170 , which converts the incoming data bit stream (Din) into clock signals with offset frequencies f0 and f1 indicative of the logic state of the bits. This induces a resonance in the RLC circuit, with the result that a magnetic field  29  is produced that wirelessly communicates the modulated data. In other words, a wireless modulated data signal  29  is produced. 
     The driving circuit  150  of  FIG. 3  has advantages and disadvantages, particularly as relates to the use of the resistor R in the circuit. One advantage of using resistor R is that it results in a wider frequency response characteristic of the RLC circuit as shown to the right in  FIG. 3 . That is, the amplitudes of the two frequencies f1 and f0 as produced in magnetic field  29  are insensitive to small variations of f1 and f0. This property of the RLC circuit makes detection at the receiver side of the communication (e.g., at the IPG  100 ) simpler: in the case the receiver&#39;s tuning does not exactly match the frequencies (f0 and f1) of the transmitter, f0 and f1 values can be adjusted by the transmitter without affecting their amplitudes to a significant degree, thus improving the likelihood of successfully detecting the communicated signal. Additionally, the resistor helps to dampen the resonance of the resonant circuit. This is important to the data rate or “bandwidth” of the communication. If resonance at one frequency (e.g., f0) can be dampened relatively quickly, a next bit of the other frequency (e.g., f1) can be accurately produced that much more quickly, which in turn allows the data rate of the communication to increase. 
     However, the resistor R also produces a significant disadvantage, namely excessive power consumption. As one skilled in the art will appreciate, a resistor dissipates energy, and hence operation of the driving circuit  150  of  FIG. 3  will need to consume more power by virtue of the passage of current through the resistor, R. In this regard, note that the driving circuit  150  is powered by the external controller  12 &#39;s rechargeable battery  76 . Because ease in use and portability suggests that the external controller  12  have an internal battery  76 , it is appreciated that this battery  76  should preferably last as long as possible, thus saving the user from the inconvenience of frequently having to recharge the battery  76 , or replace a non-rechargeable battery  76  with a fresh battery. Viewed from another perspective, excessive power consumption in the driving circuit  150  effectively limits the operable distance between the external control  12  and the IPG  100 , given that power consumption and distance are proportional. To summarize, the driving circuit  150  illustrated in  FIG. 3  can be said to have a low quality factor or “Q factor,” because the ratio of the energy in the produced field  29  to the energy used to produce that field is relatively low. 
       FIG. 4  illustrates another prior art driving circuit  150 ′, and particularly illustrates a simplified version of the approach disclosed in U.S. Pat. No. 6,349,116, which is hereby incorporated by reference in its entirety. In this circuit, a FSK modulated field  29  is also produced, but without the use of any significant resistances, R. Instead, a LC resonant circuit is formed by connecting the coil  17  and a base capacitor C 0  in parallel. This basic resonant circuit is made to resonate via control of a switch  154  which is clocked at the higher offset frequency, f1. The data modulates the frequency of the produced oscillation by switching in another capacitor, C(fsk), via switch  156 , again in parallel with L and C 0 . This has the effect of changing the resonant frequency to the lower offset frequency, f0, and thus by this arrangement, both frequencies f0 and f1 can be transmitted by either including or excluding the additional capacitance, C(fsk). 
     But just like the driving circuit of  FIG. 3 , driving circuit  150 ′ too has its advantages and disadvantages. Advantageously, because the driving circuit  150 ′ lacks a substantial resistance, the circuit is very energy efficient, and has a high Q factor. In other words, the driving circuit  150 ′ does not overtax the battery  76 , and thus allows for relatively long battery life and/or longer operating distances between the external controller  12  and the IPG  100 . 
     However, disadvantageously, driving circuit  150 ′ produces a wireless modulated data signal  29  with relatively narrow frequency response characteristic, as shown to the right in  FIG. 4 . As discussed above, this makes reception of the wireless data more difficult, and requires good matching between the transmitter and the receiver. In fact, the &#39;116 patent specifically attempts to remediate this concern by providing a bank of additional tuning capacitors, C( 1 ) through C(n). As shown in  FIG. 4 , these tuning capacitors C( 1 ) through C(n) must be selectively switched in parallel with the base capacitor C 0  using switches  158 ( 1 ) through  158 ( n ). This is regrettable, because this requires additional overhead and circuitry necessary to assess the produced frequencies and to tune them accordingly. Such added complexity makes driving circuit  150 ′ a poor choice for utilization in portable devices, and even more so for the case of implantable devices. 
     From the foregoing, it should be clear that the art of magnetically-coupled telemetry systems would benefit from a new driving circuit, one which: produces a relatively wide frequency response which allows for the reception of signals f0 and f1 without additional trimming and complexity; is able to quickly transition between logic states to enable high-speed, high-bandwidth data transfer; and is respectful of power consumption and/or can increase the operating distance between the external controller and the IPG. This disclosure provides embodiments of such a solution. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIGS. 1A and 1B  show an implantable pulse generator (IPG), and the manner in which an electrode array is coupled to the IPG in accordance with the prior art. 
         FIG. 2  shows the relation between the IPG of  FIG. 1  and an external controller with which it communicates via magnetic inductive coupling. 
         FIGS. 3 and 4  show prior art driving circuits for driving a coil to produce wireless data signals via magnetic inductive coupling. 
         FIGS. 5 and 6  show embodiments of an improved driving circuit in accordance with the invention. 
     
    
    
     DETAILED DESCRIPTION 
     The description that follows relates to use of the invention within a spinal cord stimulation (SCS) system. However, it is to be understood that the invention is not so limited. Rather, the invention may be used with any type of implantable medical device system that could benefit from improved communications between an external controller and the device. For example, the present invention may be used as part of a system employing an implantable sensor, an implantable pump, a pacemaker, a defibrillator, a cochlear stimulator, a retinal stimulator, a stimulator configured to produce coordinated limb movement, a cortical and deep brain stimulator, or in any other neural stimulator configured to treat any of a variety of conditions. 
     One embodiment of the improved driving circuit  200  is shown in  FIG. 5 , which offers significant improvement in power efficiency while at the same time being easy to implement. As will be discussed further below, the driving circuit  200  allows some of the excess current produced during resonance of the coil L to be shunted back to the rechargeable battery in the transmitter (e.g., the external controller). In so doing, the driving circuit  200  is naturally dampened, but in a way that is considerate of battery capacity. The result is a driving circuit which has a relatively broad frequency response profile. This allows frequencies f0 and f1 to be adjusted by the transmitter to match receiver characteristics without using additional circuits or hardware; is relatively responsive to allow for fast transitions between logic states and fast data rates; and is energy efficient. 
     The driving circuit  200  of  FIG. 5  uses a transformer  202  with a primary winding L 1  (which comprises the coil  17  in the external controller  12  for example) and a secondary winding L 2  in a transformer feedback configuration. As shown, no discrete resistors (such as used in the approach of  FIG. 3 ) are used in the resonant circuit, which comprises the resonance capacitor, C and the equivalent inductance L of the primary coil of the transformer  202 . The circuit  200  produces a center resonance frequency, fc, in a medical implantable system application of about 125 kHz, where fc=(2π*sqrt(LC)) −1 . 
     The primary winding L 1  ( 17 ) in conjunction with capacitor C, are made to resonate by toggling switches  160  and  160 ′. Because the switches  160  and  160 ′ are controlled with complementary clock signals, Clk and Clk′, Vbat is applied to the resonant circuit with alternating polarities. Driving the resonant circuit from both of its ends by the two switches  160  and  160 ′ has the benefit of doubling the voltage across coil L 1  and thus doubling magnetic field  29 . However, the use of two switches  160  and  160 ′ is not strictly necessary, and instead a single switch can be used in driving circuit as well (such as will be illustrated in the alternative embodiment of  FIG. 6  below). 
     The windings or coils L 1  and L 2  in the transformer  202  have N 1  and N 2  turns respectively, which set the relation of the voltages across them: VL 1 =VL 2 *N 1 /N 2 . The voltage produced on the secondary winding L 2 , VL 2 , is connected to the battery  76  via a diode D. This has the effect of limiting VL 2  to the battery voltage, Vbat. (This assumes that the threshold voltage of the diode is negligible, i.e., Vt≈0). Should VL 2  try to exceed Vbat during resonance, the diode D becomes forward biased (again, assuming that the diode has a threshold voltage of zero, i.e., Vt≈0), and a battery recovery current, i e , flows to the battery  76 . Such current flow limits the potential of VL 2  to Vbat, which in turn clamps the voltage across the coil  17  L 1  to a maximum value: VL 1 max=Vbat*(N 1 /N 2 ). (Should a significant diode threshold voltage Vt be present, this equation is modified as follows: VL 1 max=(Vbat+Vt)*(N 1 /N 2 )). By contrast, when VL 2  is less than Vbat during resonance, the diode D prevents the flow of current out of the battery  76  (i e ≈0) to prevent discharging. The bypass capacitor Cbp reduces the peak current flowing in/out of the rechargeable battery  76  to provide for stability. 
     By shunting the recovery current i e  to the rechargeable battery  76 , the battery  76  is recharged. Such recharging occurs during a portion of the time that the circuit is resonating, i.e., when the resonance produces high voltages across the coil  17  that (absent clamping) would exceed Vbat*(N 1 /N 2 ). The result is an energy efficient solution rivaling that of the solution depicted in  FIG. 4 . This is not without its drawbacks: because VL 1  is clamped to VL 1 max, the magnetic field of the wireless modulated data signal  29  produced by the coil  17  is also limited, and thus the driving circuit  200  effectively acts as a low-Q-factor circuit. However, and unlike the solution of  FIG. 3 , such magnetic field reduction does not result from energy dissipation caused through the use of a discrete resistor. 
     Moreover, shunting the recovery current i e  assists in dampening the resonance, which allows the driving circuit  200  to switch frequencies, and hence data states, more quickly, enabling the transmission of higher data rates. Such active dampening also tends to spread the width of the resonance of the driving circuit (i.e., at 121 kHz and 129 kHz). This eases the need to precisely match hardware components of the driving circuit  200  to that of the receiver, and hence allows the driving circuit to be constructed of components of lesser accuracy. 
     The various values for the components used in the improved driving circuit  200  of  FIG. 5  may be as follows for the medical implantable device system for which the circuit was originally developed. N 1 =˜22 turns; N 2 =˜3 turns; L 1 =34 μH; C=0.047 μF; Vbat=4.1V; Cbp=100 μF. However, in a different application, different component values could be chosen. 
       FIG. 6  depicts an alternative driving circuit  200 ′. Like the driving circuit  200  of  FIG. 5 , the driving circuit  200 ′ of  FIG. 6  also lacks a discrete resistor, but also allows for excess resonance energy to be shunted back to the battery  76  for the purpose of dampening and recharging. Driving circuit  200 ′ employs a single inductor L (i.e., the coil of the external controller  12  for example). A tap point  82  is added to the transmit coil  17  (in what is known in the art as an autotransformer configuration) to connect the coil  17  to the rechargeable battery  76  via a diode, D. The number of turns in the coil above (N 1 ) and below (N 2 ) the tap point  82  relates the voltage across the coil (VL) to the voltage at the tap point  82  (Vtap). Specifically, VL=Vtap*(N 2 +N 1 )/N 2 . 
     As in the transformer feedback configuration of  FIG. 5 , the diode D has the effect of clamping the voltage across the coil  17  to a maximum, VLmax. This is because the voltage at tap point  82  (Vtap) cannot exceed the battery voltage (Vbat). Should Vtap try to exceed Vbat during resonance, the diode D becomes forward biased (assuming that the diode has a threshold voltage of zero, i.e., Vt≈0), and a battery recovery current, i e , flows to the battery  76 . Such current flow limits the potential of Vtap to Vbat, which in turn clamps the voltage across the coil  17  to a maximum value: VLmax=Vbat*(N 2 +N 1 )/N 2 . (Should a significant diode threshold voltage Vt be present, this equation is modified as follows: VLmax=(Vbat+Vt)*(N 2 +N 1 )/N 2 ). By contrast, when Vtap is less than Vbat during resonance, the diode D prevents the flow of current out of the battery  76  (i e ≈0) to prevent discharging. As with the driving circuit  200  of  FIG. 5 , the center frequency of resonance is fc=(2π*sqrt(LC)) −1 , where L comprises the inductance of the coil L  17 . 
     The effect, as with the driving circuit  200  of  FIG. 5 , is that recovery current i e  is shunted to the rechargeable battery  76  during high voltage portions of the resonance, providing a very energy efficient solution. Although VL is clamped to VLmax, which limits the magnetic field producible by the coil  17 , such magnetic field reduction does not result from energy dissipation caused through the use of a discrete resistor. And once again, shunting the recovery current i e  to the battery assists in dampening the resonance, which allows for the transmission of higher data rates. Additionally, and as before, such active dampening also tends to widen the resonant response of the resonant circuit, easing reception without the need for hardware trimming. 
     The various values for the components used in the improved driving circuit  200 ′ of  FIG. 6  may be as follows for the medical implantable device system for which the circuit was originally developed. N 1 =˜20 turns; N 2 =˜4 turns; L=34 μH; C=0.047 μF; Vbat=4.1V; Cbp=100 μF. However, in a different application, different component values could be chosen. 
     The disclosed driving circuits can be used with any switching type amplifier (Class C, D, E, H, etc.), and for both the transformer feedback configuration ( FIG. 5 ) and the autotransformer ( FIG. 6 ) configuration. 
     While disclosed in the context of a medical implantable device system for which the invention was originally contemplated, it should be recognized that the improved driving circuitry disclosed herein is not so limited, and can be used in other contexts employing communications via magnetic inductive coupling, such as in Radio-Frequency Identification (RFID) systems, etc. The disclosed circuitry can further be used in any context in which magnetic inductive coupling could be used as a means of communication, even if not so used before. 
     Although particular embodiments of the present invention have been shown and described, it should be understood that the above discussion is not intended to limit the present invention to these embodiments. It will be obvious to those skilled in the art that various changes and modifications may be made without departing from the spirit and scope of the present invention. Thus, the present invention is intended to cover alternatives, modifications, and equivalents that may fall within the spirit and scope of the present invention as defined by the claims.