Abstract:
A frequency synthesizer within an FM receiver employs a Phase-Locked Loop (PLL) to generate a Local Oscillator (LO) signal. The LO signal is supplied to a mixer. The FM receiver also includes jammer detection functionality. If no jammer is detected, then the loop bandwidth of the PLL is set to have a relatively high value, thereby favoring suppression of in-band residual FM. If a jammer is detected, then the loop bandwidth of the PLL is set to have a relatively low value, thereby favoring suppression of out-of-band SSB phase noise. By adaptively changing loop bandwidth depending on whether a jammer is detected, performance requirements on sub-circuits within the PLL can be relaxed while still satisfying in-band residual FM and out-of-band SSB phase noise requirements. By allowing the VCO of the PLL to generate more phase noise due to the adaptive changing of loop bandwidth, VCO power consumption can be reduced.

Description:
BACKGROUND INFORMATION 
       [0001]    1. Technical Field 
         [0002]    The present disclosure relates to Phase-Locked Loop (PLL) based frequency synthesizers within Frequency Modulation (FM) receivers. 
         [0003]    2. Background Information 
         [0004]    A miniaturized and integrated FM (Frequency Modulation) radio transceiver generally involves the use of a frequency synthesizer. The frequency synthesizer typically includes a Phase-Locked Loop (PLL) and therefore may be referred to as a PLL-based frequency synthesizer. If, for example, an integrated FM radio transceiver is receiving a radio transmission, then the PLL-based frequency synthesizer within the transceiver generates a Local Oscillator (LO) signal. The LO signal is supplied to a mixer that is part of a demodulator of an FM receiver portion of the transceiver. If, on the other hand, the FM radio transceiver is transmitting a radio transmission, then the same PLL-based frequency synthesizer is used to output an FM modulated signal. The FM modulated signal is then amplified and supplied to the antenna for transmission. Of various requirements imposed on the design of such a PLL-based frequency synthesizer within an FM receiver, there are two requirements of particular importance: 1) In-band residual FM, and 2) Out-of-band SSB (Single Side Band) phase noise. 
         [0005]    Table 1 below sets forth an example of requirements that may be imposed on a LO signal output by a PLL-based frequency synthesizer within an FM transceiver. 
         [0000]    
       
         
               
               
               
               
             
               
               
               
               
             
           
               
                 TABLE 1 
               
               
                   
               
               
                 Specification 
                 Comments 
                 Max 
                 Units 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 In-Band Residual FM Mono 
                 300 Hz-5 KHz 
                 19 
                 Hzrms 
               
               
                 In-Band Residual FM Stereo L-R 
                 33 KHz-43 KHz 
                 67 
                 Hzrms 
               
               
                 In-Band Residual FM Stereo RDS 
                 55 KHz-59 KHz 
                 67 
                 Hzrms 
               
               
                 Out-of-Band SSB Phase Noise 
                 at 200 KHz 
                 −112 
                 dBc/Hz 
               
               
                   
                 at 400 KHz 
                 −124 
                 dBc/Hz 
               
               
                   
                 at &gt;500 KHz 
                 −126 
                 dBc/Hz 
               
               
                   
               
             
          
         
       
     
         [0006]      FIG. 1  (Prior Art) is a diagram that shows a 59 KHz wide portion of the FM band above an FM carrier frequency. For each FM broadcast radio station, there is such a 59 KHz wide portion above the FM carrier, and one such 59 KHz wide portion below the FM carrier. Generally speaking, in-band residual FM is found by integrating phase noise in the LO signal over one of the particular frequency range portions of  FIG. 1 . For example, in-band residual FM monotone is a measure of SSB phase noise in the monotone frequency range 1 of  FIG. 1 . In-band residual FM monotone (300 Hz to 5 KHz) can be determined by integrating SSB (Single Side Band) phase noise over the 300 Hz to 5 KHz range in accordance with Equation (1) below, where L(f) is measured PLL phase noise and has units of dBc/Hz: 
         [0000]    
       
         
           
             
               
                 
                   
                     FMres 
                     Mono 
                   
                   = 
                   
                     
                       2 
                       · 
                       
                         
                           ∫ 
                           
                             300 
                              
                             
                                 
                             
                              
                             Hz 
                           
                           
                             5000 
                              
                             
                                 
                             
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                             Hz 
                           
                         
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                               2 
                             
                             · 
                             
                               10 
                               
                                 
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                                    
                                   
                                     ( 
                                     f 
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         [0007]    In-band residual FM stereo is a measure of SSB phase noise in the stereo frequency range 2 of  FIG. 1 . In-band residual FM stereo can be determined by integrating SSB phase noise over the 33 KHz to 43 KHz frequency range in accordance with Equation (2) below: 
         [0000]    
       
         
           
             
               
                 
                   
                     FMres 
                     Stereo_LR 
                   
                   = 
                   
                     
                       2 
                       · 
                       
                         
                           ∫ 
                           
                             33000 
                              
                             
                                 
                             
                              
                             Hz 
                           
                           
                             43000 
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                     2 
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         [0008]    In-band residual FM RDS (Radio Data System) is a measure of phase noise in the RDS/RBDS frequency range 3 of  FIG. 1 . In-band residual FM RDS can be determined by integrating SSB phase noise over the 55 KHz to 59 KHz frequency range in accordance with Equation (3) below: 
         [0000]    
       
         
           
             
               
                 
                   
                     FMres 
                     RDS 
                   
                   = 
                   
                     
                       2 
                       · 
                       
                         
                           ∫ 
                           
                             55000 
                              
                             
                                 
                             
                              
                             Hz 
                           
                           
                             59000 
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                             Hz 
                           
                         
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         [0009]    The quality of audio output by an FM receiver is generally limited by the in-band residual FM. When no jammer is present, residual FM is usually the limiting performance parameter and determines the effective audio Signal-to-Noise Ratio (SNR) and hence audio quality provided that the Carrier-to-Noise Ratio (CNR) is high. When a jammer is present, however, then the quality of the audio output by the FM receiver is generally limited by out-of-band SSB phase noise. The jammer and the wanted signal being received are FM demodulated, giving rise to audio distortion due to reciprocal mixing. This audio distortion dominates the deleterious effect of the in-band residual FM phase noise. In one example, a jammer is a signal that is of such a frequency and power that it reciprocally mixes with phase noise of the LO signal in the receiver in such a way that substantial in-band signal-to-noise degradation results. The transmitted FM signal of an adjacent FM radio channel may be an example of one such jammer. 
         [0010]    In a conventional miniaturized and integrated FM receiver involving a PLL-based frequency synthesizer, the various components of the PLL-based frequency synthesizer are generally sized and tuned to achieve acceptable performance under both the no jammer condition and the jammer condition. 
       SUMMARY 
       [0011]    A frequency synthesizer within an FM receiver employs a Phase-Locked Loop (PLL) to generate a Local Oscillator (LO) signal. The LO signal is supplied to a mixer in the process of demodulating an FM signal. The FM receiver also includes jammer detection functionality. In accordance with one novel aspect, the loop bandwidth of the PLL is changed based at least in part on whether the jammer detection functionality detects a jammer. 
         [0012]    In one specific example, if no jammer is detected by the jammer detection functionality, then the loop bandwidth of the PLL is set to have a relatively high loop bandwidth. The relatively high PLL loop bandwidth favors reducing in-band residual FM. If, however, a jammer is detected by the jammer detection functionality, then the loop bandwidth of the PLL is set to have a relatively low loop bandwidth. The relatively low PLL loop bandwidth favors suppression of out-of-band SSB phase noise. By automatically and adaptively changing the loop bandwidth of the PLL depending on whether a jammer is detected, performance requirements on sub-circuits of the PLL are relaxed while still satisfying in-band residual FM requirements and out-of-band SSB phase noise requirements. For example, achieving low Voltage Controlled Oscillator (VCO) phase noise often requires that the VCO sub-circuit within the PLL consume a relatively large amount of power or that a large layout area high quality factor spiral inductor be used in the LC tank of the VCO. By allowing the VCO sub-circuit to generate more phase noise due to the adaptive changing of PLL loop bandwidth described above, the amount of power consumed by the VCO in an FM receiver can be reduced while still satisfying performance requirements imposed on the FM receiver. Similarly, by allowing the VCO sub-circuit to generate more phase noise due the adaptive changing of PLL loop bandwidth, the size of the PLL can be smaller due to not having to use a large layout area high quality factor spiral inductor. 
         [0013]    The foregoing is a summary and thus contains, by necessity, simplifications, generalizations and omissions of detail; consequently, those skilled in the art will appreciate that the summary is illustrative only and does not purport to be limiting in any way. Other aspects, inventive features, and advantages of the devices and/or processes described herein, as defined solely by the claims, will become apparent in the non-limiting detailed description set forth herein. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0014]      FIG. 1  (Prior Art) is a diagram that shows a 59 KHz wide frequency range relative to a FM carrier frequency. 
           [0015]      FIG. 2  is a simplified diagram of a mobile communication device  100  in accordance with one novel aspect. 
           [0016]      FIG. 3  is a more detailed diagram of the FM transmitter/receiver integrated circuit  108  of the mobile communication device of  FIG. 2 . 
           [0017]      FIG. 4  is a diagram that shows how the VCO phase noise contribution to total PLL phase noise varies as a function of frequency. 
           [0018]      FIG. 5  is a diagram that shows how the charge pump phase noise contribution to total PLL phase noise varies as a function of frequency. 
           [0019]      FIG. 6  is a diagram that shows how the loop filter phase noise contribution to total PLL phase noise varies as a function of frequency. 
           [0020]      FIG. 7  is a diagram of the jammer detection functionality  148  of  FIG. 3 . 
           [0021]      FIG. 8  is a flowchart of a method  200  of adjusting PLL loop bandwidth based on jammer detection information. If no jammer is detected then a relatively high PLL loop bandwidth is used, whereas if a jammer is detected then a relatively low PLL loop bandwidth is used. 
           [0022]      FIG. 9  is a more detailed diagram of the loop filter  140  and the charge pump  139  in the PLL  136  of the FM transmitter/receiver integrated circuit  108  of  FIG. 3 . 
           [0023]      FIG. 10  is a table that sets forth the various possible settings of the circuit elements of loop filter  140  and charge pump  139  of  FIG. 9 . 
           [0024]      FIG. 11  is a table that sets forth how the circuit elements of loop filter  140  and charge pump  139  should be set to configure the PLL to have a relatively high PLL loop bandwidth setting, and to configure the PLL to have a relatively low PLL bandwidth setting. 
           [0025]      FIG. 12  is a graph showing how the total phase noise of the PLL  136  of  FIG. 3  varies over frequency in an operational condition in which no jammer is detected and in which the PLL loop bandwidth has its first relatively high value of 183 KHz in accordance with the method of  FIG. 8 . 
           [0026]      FIG. 13  is a graph showing how the total phase noise of the PLL  136  of  FIG. 3  varies over frequency in an operational condition in which a jammer is detected and in which the PLL loop bandwidth has its second relatively low value of 125 KHz in accordance with the method of  FIG. 8 . 
       
    
    
     DETAILED DESCRIPTION 
       [0027]      FIG. 2  is a very simplified high level block diagram of one particular type of mobile communication device  100  that carries out a Phase-Locked Loop (PLL) bandwidth adjustment method in accordance with one aspect. In the present example, mobile communication device  100  is a battery-powered handheld device such as a cellular telephone. Cellular telephone  100  includes (among other parts not illustrated) an antenna  101  usable for receiving and transmitting cellular telephone communications, an RF transceiver integrated circuit  102 , and a digital baseband integrated circuit  103 . In one very simplified explanation of the operation of the cellular telephone, if the cellular telephone is being used to receive audio information as part of a cellular telephone conversation, then an incoming transmission  104  is received on antenna  101 . The signal is amplified and downconverted and filtered in RF transceiver integrated circuit  102 . After being digitized, demodulated and decoded in digital baseband integrated circuit  103 , the resulting audio information  105  may, for example, be used to drive a speaker (not shown) such that a user of the cellular telephone can hear another speaker in the cellular telephone conversation. If, on the other hand, cellular telephone  100  is to be used to transmit to audio information as part of the cellular telephone conversation, then a microphone (not shown) that is part of the mobile communication device receives sound and converts that sound into an electrical signal. The electrical signal is converted into a stream of digital values of audio information  106 . Audio information  106  is encoded, modulated and converted into analog form in digital baseband integrated circuit  103 . The resulting analog signal is filtered and upconverted in RF transceiver integrated circuit  102 . After being amplified, the signal is transmitted from antenna  101  as transmission  107 . This explanation of cellular telephone operation is very simplified and is presented here to provide a context for explaining operation of the PLL bandwidth adjustment method. 
         [0028]    In addition to the cellular telephone functionality described above, cellular telephone  100  has an ability to receive and to transmit FM radio communications (FM VHF broadcast band communications from approximately 76 MHz to approximately 108 MHz). To provide this FM microtransmitter radio functionality, cellular telephone  100  includes a FM transmitter/receiver integrated circuit  108  that is coupled to digital baseband integrated circuit  103  via a serial bus  109 . A user may, for example, use cellular telephone  100  to receive and to listen to ordinary FM broadcast radio stations in the FM VHF band. When cellular telephone  100  is used in this way, an FM radio signal  110  is received onto a printed circuit board (PCB) antenna  111 , and is supplied via a matching network  119  to FM transmitter/receiver integrated circuit  108 . In the alternative, if a headset  113  is attached to the cellular telephone via a connector  114 , then FM radio signal  110  is received onto antenna  113  and is supplied via matching network  115  to FM transmitter/receiver integrated circuit  108 . The incoming FM signal is demodulated by an FM receiver functionality  116 . The resulting information received can then be communicated via serial bus  109  to digital baseband integrated circuit  103 . Digital baseband integrated circuit  103  can then drive the speaker or headset of the user such that the user can listen to the FM broadcast information. In this way, a user of cellular telephone  100  can use cellular telephone  100  to listen to ordinary FM radio stations in the 76 MHz to 108 MHz FM band. 
         [0029]    Cellular telephone  100  may also be used to transmit FM signals in the same FM VHF band. A user may, for example, use the audio system of an automobile or of a home stereo system to listen to audio information stored on the cellular telephone. In one example, an audio file such as an MP3 file is stored on cellular telephone  100  and the user wishes to hear audio of the file on the sound system of the user&#39;s automobile. To do this, the MP3 file is communicated from digital baseband integrated circuit  103  via serial bus  109  to FM transmitter/receiver integrated circuit  108 . The MP3 information is converted into a stream of audio information that is then FM modulated onto a carrier by an FM transmitter functionality  117 . An FM radio signal is then driven onto antenna  111  or onto antenna  113  if it is provided. The resulting FM transmission  118  may then be received by the FM radio tuner in the user&#39;s automobile. The FM radio of the automobile then receives the FM transmission  118  and drives the speakers in the automobile as it would if it were tuned to receive an ordinary FM radio station. In this way, the user can use cellular telephone  100  to play MP3 music in the user&#39;s automobile where the MP3 music is stored in cellular telephone  110 . This can be accomplished without connecting any wires between cellular telephone  100  and the FM radio system of the automobile. 
         [0030]      FIG. 3  is a more detailed diagram of the FM transmitter/receiver integrated circuit  108  of  FIG. 2 . The FM receive path extends from PCB antenna  111  or from headset wire antenna  113 , through a transmit/receive (TR) front end switch  120 , through a low noise amplifier (LNA)  121 , through a mixer block  122 , through a complex bandpass filter  123 , through a pair of analog-to-digital converters (ADCs)  124  and  125 , and to a Digital Signal Processor (DSP)  126 . A local oscillator signal (LO)  128  generated by a frequency synthesizer  129  is supplied to the mixer block  122 . Arrow  127  represents the resulting stream of digitized audio information. 
         [0031]    The FM transmit path extends from conductors  130 . Arrow  131  represents the incoming stream of digitized audio information. A DSP  132  and an associated sigma-delta modulator  133  operate together to supply a stream  134  of digital values to the frequency synthesizer  129 . This stream of digital values  134  causes the frequency synthesizer  129  to output an FM signal  135 . FM signal  135  is buffered by buffer  137 A and is amplified by a power amplifier (PA)  137 . FM signal  135  then passes through TR switch  120 , and to an antenna ( 111  and/or  113 ) for transmission. Accordingly, the same frequency synthesizer  129  is used in both the receive path and in the transmit path. 
         [0032]    Frequency synthesizer  129  includes a fractional-N Phase-Locked Loop (PLL) portion  136  and a programmable output divider portion  137 . PLL portion  136  includes a Phase-Frequency Detector (PFD)  138 , a charge pump  139 , a loop filter  140 , a Voltage-Controlled Oscillator (VCO)  141 , a VCO buffer  141 A, and a loop divider  142 . A 19.2 MHz reference clock signal  143  is supplied from an external reference (for example, from an external crystal oscillator). PFD  138  compares the phase of a feedback signal  144  to the phase of the reference clock signal  143 , and controls charge pump  139  appropriately such that a DC control signal  145  supplied to VCO  141  is increased or decreased. The DC control signal  145  is increased or decreased such that the phase of feedback signal  144  is remains locked to the phase of the reference clock signal  143 . Arrow  59  identifies the VCO output signal after buffering by buffer  141 A. 
         [0033]    In an FM receiver such as the FM receiver of  FIG. 3 , limits are placed on the amount of permissible phase noise in the local oscillator signal LO  128 . Phase noise is a quantity that indicates the degree of spectral purity of the LO signal. Equation 4 below is an equation for total phase noise in the local oscillator signal LO  128 . 
         [0000]    
       
         
           
             
               
                 
                   
                     
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         [0034]    As indicated by Equation 4, total phase noise includes several components including a component S θCP  due to charge pump  139 , a component S θTdiv  due to loop divider  142 , a component S θVCO  due to VCO  141 , a component S θFLT  due to loop filter  140 , a component S θREF  due to noise in the input reference clock signal  143 , a component S θBUFF  due to the VCO output buffer  141 A, and a component S θΣΔ  due to sigma delta modulator  133 . 
         [0035]      FIG. 4  is a diagram that shows how the VCO phase noise contribution to total phase noise varies as a function of frequency. 
         [0036]      FIG. 5  is a diagram that shows how the charge pump phase noise contribution to total phase noise varies as a function of frequency. 
         [0037]      FIG. 6  is a diagram that shows how the loop filter phase noise contribution to total phase noise varies as a function of frequency. 
         [0038]    In one novel aspect, residual FM phase noise requirements imposed on the FM receiver are considered and SSB phase noise requirements imposed on the FM receiver are considered. First, it is recognized that if no jammer is present, then the residual FM phase noise requirements are generally more difficult to meet than are the out-of-band SSB phase noise requirements. Audio quality is generally limited in a conventional FM receiver due to in-band residual FM approaching residual FM phase noise requirements, whereas SSB phase noise requirements are easily satisfied. Increasing PLL loop bandwidth generally serves to improve residual FM (decrease residual FM) but unfortunately also serves to degrade out-of-band SSB phase noise (increase out-of-band SSB phase noise). 
         [0039]    Second, it is further recognized that if a jammer is present, then the out-of-band SSB phase noise requirements are generally more stringent than are the residual FM phase noise requirements. Audio quality is generally limited in a conventional FM receiver due to out-of-band SSB phase noise approaching the out-of-band SSB phase noise requirements, whereas the residual FM phase noise requirements are generally satisfied by some margin. Decreasing PLL loop bandwidth generally serves to improve out-of-band SSB phase noise (decrease SSB phase noise) but unfortunately also serves to degrade residual FM phase noise (increase residual FM phase noise). 
         [0040]    It is further recognized that the presence of a jammer can often be detected. This detection is performed in software or firmware by a jammer detection functionality  148  in DSP  126  of  FIG. 3 . The software or firmware program of processor-executable instructions (also referred to a computer-executable instructions) is stored in a processor-readable medium (also referred to as a computer-readable medium) that is in or is coupled to DSP  126 . The instructions are executed by DSP  126 . 
         [0041]      FIG. 7  is a more detailed functional diagram of one example of jammer detection functionality  148 . The presence of a jammer is detected by comparing the total power I of the incoming received signal before a digital filtering operation (RSSI) with the total power C of the incoming received signal after the digital filtering operation (RMSSI). Function block  155  in  FIG. 7  represents the digital filtering operation. RSSI stands for Received Signal Strength Indicator. The RSSI indicator is generated by function block  156  of  FIG. 7 . RMSSI stands for Received Mean Strength Signal Indicator. The RMSSI indicator is generated by function block  157  of  FIG. 7 . Comparator block  157  compares the RSSI and RMSSI indicators and generates an IX ratio  149 . The I/C ratio indicates both the presence of a jammer, as well as the strength of the jammer. I/C ratio  149  is an indication of whether a jammer is present or not. This determination is but one way that the presence of a jammer can be detected. Any other suitable way of detecting a jammer can be employed. 
         [0042]      FIG. 8  is a flowchart of a jammer detection based PLL bandwidth adjustment method  200  in accordance with one novel aspect. If a jammer is not detected (step  201 ), then PLL  136  is controlled to have a first PLL loop bandwidth (step  202 ). The first PLL loop bandwidth is relatively high in order to favor and facilitate suppression of in-band residual FM. In one example, the FM receiver of integrated circuit  108  is simulated or tested in operating conditions in which no jammer is present, and the optimum PLL loop bandwidth that results in the best audio quality is recorded, considering all requirements including residual FM requirements and out-of-band SSB phase noise requirements. This determined PLL bandwidth is the first PLL loop bandwidth. In one specific example, this first bandwidth is 183 KHz. 
         [0043]    If, however, a jammer is detected (step  201 ), then PLL  136  is controlled to have a second PLL loop bandwidth (step  203 ). The second PLL loop bandwidth is relatively low (as compared to the first PLL bandwidth) in order to favor suppression of out-of-band SSB phase noise. In one example, the FM receiver of integrated circuit  108  is simulated or tested in operating conditions in which a jammer is present, and the optimum PLL loop bandwidth that results in the best audio quality is recorded, considering all requirements including residual FM requirements and out-of-band SSB phase noise requirements. This determined PLL loop bandwidth is the second PLL loop bandwidth. In one specific example, this second bandwidth is 125 KHz. 
         [0044]    To facilitate changing the loop bandwidth of PLL  136 , loop filter  140  is a programmable loop filter.  FIG. 9  is a more detailed diagram of programmable loop filter  140 . Programmable loop filter  140  includes resistance elements R 1  and R 2  that have digitally-controlled variable resistances. Programmable loop filter  140  also includes capacitance elements C 1 , C 2  and C 3  that have digitally-controlled capacitances. The resistances and capacitances of the resistance elements and capacitance elements are determined by a multi-bit digital loop filter control value on conductors  150  of  FIG. 3 . To facilitate changing the loop bandwidth of PLL  136 , charge pump  139  is also a programmable charge pump. If the one-bit digital current control value on conductor  151  of  FIG. 3  has a first digital value, then charge pump  139  is controlled to sink and source a first amount of current I CP , whereas if the one-bit digital current control value on conductor  151  has a second digital value, then charge pump  139  is controlled to sink and source a second amount of current I CP . Although not utilized in the particular operational example described here, the tuning sensitivity (Kvco) of VCO  141  is variable and can be set to have one of two values as determined by a one-bit digital K VCO  control value supplied to VCO  141  via conductor  152 . In the presently described operational example, the tuning sensitivity of VCO  141  is set to have a constant value and the one-bit digital control value on conductor  152  is not changed. PLL bandwidth control logic functionality  153  within DSP  126  generates the digital control values on conductors  150 ,  151 , and  152  based at least in part on whether or not a jammer has been detected as determined by jammer detection functionality  148 . The control values on conductors  150 ,  151  and  152  together form a multi-bit control signal  154 . Upon detection of a jammer by jammer detection functionality  148 , the value of the multi-bit control signal  154  changes from a first value to a second value such that the bandwidth of the PLL changes from the first PLL loop bandwidth to the second PLL loop bandwidth. If after a time the jammer detection functionality  148  no longer detects the jammer, then the value of multi-bit control signal  154  changes from the second value to the first value such that the PLL loop bandwidth is changed back from the second PLL loop bandwidth to the first PLL loop bandwidth. 
         [0045]      FIG. 10  is a table that sets forth the different resistance, capacitance, and current values that the various resistance elements, capacitance elements, and current source elements of  FIG. 9  can be controlled to have. For example, capacitance element C 1  of  FIG. 9  can be set to have a selectable one of sixteen capacitances in a range from 25 picofarads to 220 picofarads. A four-bit digital value determines which of these sixteen capacitances the capacitor C 1  will have. Similarly, the resistance of resistance element R 1  is determined by another four-bit digital value. The capacitance of capacitance element C 2  is determined by a three-bit digital value. The resistance of resistance element R 2  is determined by a one-bit digital value. All these digital values are supplied together to loop filter  140  via conductors  150 . Similarly, the charge pump current I CP  can be set to have a value of 39 microamperes, or a value of 85 microamperes. Which of these two current values it is that charge pump  139  will sink and source is determined by the one-bit digital current control value supplied via conductor  151  to charge pump  139 . 
         [0046]      FIG. 11  is a table that sets forth one particular example of how PLL  136  of  FIG. 3  is configured in step  202  (in method  200  of  FIG. 8 ) to have a first relatively high PLL loop bandwidth, and how PLL  136  is configured in step  203  to have the second relatively low PLL loop bandwidth. Column  300  sets forth how the values of circuit elements R 1 , R 2 , C 1 , C 2  and C 3  and the charge pump current I CP  are set such that the PLL loop bandwidth has its first bandwidth value of approximately 183 KHz. As indicated in the flowchart of  FIG. 8 , these settings are used if no jammer is detected. This first high PLL loop bandwidth results in optimal audio quality in operating conditions involving no jammer. Column  301 , on the other hand, sets forth how the values of circuit elements R 1 , R 2 , C 1 , C 2  and C 3  and the charge pump current I CP  are set such that the PLL loop bandwidth has its second relatively low bandwidth value of approximately 125 KHz. As indicated in the flowchart of  FIG. 8 , these settings are used if a jammer is detected. The low PLL loop bandwidth results in optimal audio quality in operating conditions involving a jammer. The capacitances of the capacitance elements C 1 , C 2  and C 3  are not changed when switching between the two PLL loop bandwidth configurations, thereby minimizing disturbances to the PLL and reducing PLL settling time. 
         [0047]      FIG. 12  is a graph showing how the total phase noise  400  of PLL  136  of  FIG. 3  varies over frequency in an operational condition in which no jammer is detected and in which the PLL loop bandwidth has its first relatively high value of 183 KHz. Total phase noise  400  is below the receiver mask  408  at all frequencies. 
         [0048]      FIG. 13  is a graph showing how the total phase noise  500  of PLL  136  of  FIG. 3  varies over frequency in an operational condition in which a jammer is detected and in which the PLL loop bandwidth has its second relatively low value of 125 KHz. Total phase noise  500  is below the receiver mask  508  at all frequencies. 
         [0049]    In one or more exemplary embodiments, the functions described may be implemented in hardware, software, firmware, or any combination thereof. If implemented in software, the functions may be stored on or transmitted over as one or more instructions or code on a computer-readable medium. Computer-readable media includes both computer storage media and communication media including any medium that facilitates transfer of a computer program from one place to another. A storage media may be any available media that can be accessed by a general purpose or special purpose computer. By way of example, and not limitation, such computer-readable media can comprise RAM, ROM, EEPROM, CD-ROM or other optical disk storage, magnetic disk storage or other magnetic storage devices, or any other medium that can be used to carry or store desired program code means in the form of instructions or data structures and that can be accessed by a general-purpose or special-purpose computer, or a general-purpose or special-purpose processor. Also, any connection is properly termed a computer-readable medium. For example, if the software is transmitted from a website, server, or other remote source using a coaxial cable, fiber optic cable, twisted pair, digital subscriber line (DSL), or wireless technologies such as infrared, radio, and microwave, then the coaxial cable, fiber optic cable, twisted pair, DSL, or wireless technologies such as infrared, radio, and microwave are included in the definition of medium. Disk and disc, as used herein, includes compact disc (CD), laser disc, optical disc, digital versatile disc (DVD), floppy disk and blu-ray disc where disks usually reproduce data magnetically, while discs reproduce data optically with lasers. Combinations of the above should also be included within the scope of computer-readable media. 
         [0050]    Although certain specific embodiments are described above for instructional purposes, the teachings of this patent document have general applicability and are not limited to the specific embodiments described above. Jammer detection can be carried out in software as illustrated in  FIG. 3 , or can be carried out by hardware circuitry. Jammer detection can occur in the same integrated circuit that contains the PLL, or can occur in another integrated circuit that does not contain the PLL. The ways of changing PLL loop bandwidth set forth above are but examples. Other PLL circuit components can be controlled to change PLL loop bandwidth, and the PLL circuit components that are described as being controlled in the description above can be controlled in other ways to change PLL loop bandwidth. The determination to change PLL loop bandwidth need not depend entirely on whether or not a jammer has been detected, but rather may also depend in part on other information. Although the jammer detection based adaptive PLL bandwidth adjustment method is described above in connection with a cellular telephone example involving an FM microtransmitter, the method sees general applicability in FM receivers. Accordingly, various modifications, adaptations, and combinations of the various features of the described specific embodiments can be practiced without departing from the scope of the claims that are set forth below.