Abstract:
In a logic circuit having PMOS pull-up devices and NMOS pull-down devices, the PMOS pull-up devices are sized relative to the NMOS pull-down devices according to the number of transistors that simultaneously turn on. In one embodiment, the PMOS transistor width is determined by multiplying the effective NMOS transistor width by a predetermined factor indicative of a current carrying ratio between one of the PMOS pull-up transistors and one of the NMOS pull-down transistors and dividing by the number of PMOS pull-up transistors that simultaneously turn on to charge the output node high. Where the PMOS pull-up devices are parallel-connected, the NMOS transistor width is divided by the number of NMOS transistors.

Description:
BACKGROUND 
     1. Field of Invention 
     This invention relates generally to CMOS logic and specifically to minimizing silicon area of PMOS pull-up devices in static logic. 
     2. Description of Related Art 
     CMOS logic devices typically include one or more PMOS pull-up transistors connected between a voltage supply and an output node and one or more NMOS pull-down transistors connected between the output node and ground potential. Since the mobility of electrons is greater than the mobility of holes, an NMOS transistor is able to conduct a greater current than a PMOS transistor of the same size. Consequently, in CMOS logic devices, the PMOS pull-up transistors are typically sized much larger than their corresponding NMOS pull-down transistors so that the charge path formed by the PMOS pull-up transistors and the discharge path formed by the NMOS pull-down transistors have equal drive strengths. Maintaining equal drive strengths for the charge and discharge paths is necessary to achieve equal charging and discharging rates of the output node, which in turn provides balanced logic transitions. 
     A well-known sizing factor used to maintain equal drive strength between a PMOS pull-up transistor and an NMOS pull-down transistor is Beta (β), which is equal the mobility of electrons divided by the mobility of holes. This relationship may be expressed as β≅μ n /μ p , where μ n  is the mobility of electrons, and μ p  is the mobility of holes. Since the effective drive strength of a transistor is proportional to its width W, the width W p  of a PMOS transistor which provides the same drive strength as an NMOS transistor of width W n  is given by W p =βW n . 
     FIG. 1 shows a conventional CMOS inverter  10  having a PMOS pull-up transistor  11  connected between a supply voltage V DD  and an output node  12 , and having an NMOS pull-down transistor  13  connected between the output node  12  and ground potential. The CMOS inverter inverts an input signal provided to the gates of transistors  11  and  13  to generate an output signal B at output node  12 . When signal A is logic low, the PMOS pull-up transistor turns on and charges the output node  12  toward V DD , while the NMOS pull-down transistor  13  turns off and isolates output node  12  from ground potential. When signal A is logic high, the NMOS pull-down transistor  13  turns on and discharges output node  12  toward ground potential, while the PMOS pull-up transistor  11  turns off and isolates output node  12  from V DD . Since the PMOS transistor  11  is the only pull-up device, the effective drive strength S p  of the pull-up path is determined by the width W p  of the pull-up transistor  11 . Similarly, since the NMOS transistor  13  is the only pull-down device, the effective drive strength of the pull-down path is determined by the width W n  of the pull-down transistor  13 . Thus, to maintain equal drive strengths S p  and Sn for the respective charge and discharge paths in the CMOS inverter  10 , the width of the PMOS pull-up transistor W p  should be equal to the width of the NMOS pull-down transistor W n  times Beta, i.e., W p =βW n . For example, if β=2, the width W p  of the PMOS pull-up transistor  11  must be twice the width W n  of the NMOS pull-down transistor  13  to achieve equal drive strengths for the charge and discharge paths. 
     FIG. 2 shows a conventional NAND gate  20  having two PMOS pull-up transistors  21  and  22  connected in parallel between V DD  and an output node  23 , and two NMOS pull-down transistors  24  and  25  connected in series between the output node  23  and ground potential. A first input signal A 0  is provided to the respective gates of the transistors  21  and  24 , and a second input signal Al is provided to the respective gates of transistors  22  and  25 . If either of the input signals A 0  or A 1  is logic low, the corresponding PMOS pull-up transistor  21  and/or  22  turns on and charges the output node  23  toward V DD , thereby driving the output signal B to logic high. Since only one of the input signals A 0  or A 1  must be logic low to drive output signal B to logic high, sometimes only one of the PMOS pull-up transistors  21  and  22  turns on to charge the output node  23  toward V DD . Thus, the effective drive strength S p  of the charge path is equal to that of one of the PMOS pull-up transistors  21  or  21 , i.e., S p ≅W p . 
     Both input signals A 0  and A 1  must be logic high in order to discharge the output node  23  toward ground potential through series-connected NMOS pull-down transistors  24  and  25 . Since the resistance of two series-connected transistors is twice that of a single transistor, the effective drive strength of the discharge path through NMOS pull-down transistors  24  and  25  is about one-half the effective drive strength of a single NMOS transistor, i.e., S n ≅W n /2. Thus, in order to maintain equal drive strengths for the charge and discharge paths of the NAND gate  20 , the width of each of the PMOS pull-up transistors  21  and  22  is sized by multiplying the effective NMOS transistor width times Beta, i.e., W p =βW n /2. 
     It is always desirable to reduce the size of a circuit such as, for instance, a CMOS logic device, since any reduction in circuit size typically reduces manufacturing costs and power consumption. Further, reducing the silicon area of a circuit advantageously allows for the circuit to be more easily fabricated using smaller technologies. 
     While it is desirable to reduce transistor size in order to conserve silicon area, the proper ratio between PMOS and NMOS devices in logic devices must be maintained in order to preserve the balance of the circuit. Otherwise, one input level may overpower the other input level, which in turn could result in erroneous data. Thus, it is desirable to reduce transistor size without sacrificing performance or upsetting the balance of current-carrying capability between PMOS pull-up and NMOS pull-down devices. 
     SUMMARY 
     A method is disclosed that allows for a reduction in silicon area in applications where a number of input signals simultaneously transition to the same logic state. In accordance with one embodiment of the present invention, the PMOS pull-up devices of a logic circuit are sized relative to the NMOS pull-down devices therein according to the number of the PMOS pull-up devices that simultaneously turn on. For example, in one embodiment, the PMOS pull-up transistor width is determined by multiplying the NMOS pull-down transistor width by a predetermined factor indicative of a current carrying ratio between one of the PMOS pull-up transistors and one of the NMOS pull-down transistors and then dividing by the number of PMOS pull-up transistors that simultaneously turn on to charge the output node high. In one embodiment, the width W p  of each parallel-connected PMOS pull-up device sized relative to the width W n  of each series-connected NMOS pull-down device may be expressed as W p =W n (β/kN), where β is the current carrying ratio between one of the PMOS pull-up transistors and one of the NMOS pull-down transistors, k is the number of PMOS devices that simultaneously turn on, and N is the number of NMOS devices. In another embodiment, the width W p  of each series-connected PMOS pull-up device sized relative to the width W n  of each parallel-connected NMOS pull-down device may be expressed as W p =W n (βN/k). By factoring the number of PMOS pull-up devices that simultaneously turn on into sizing of the PMOS pull-up devices, the present invention may reduce silicon area of logic devices having PMOS pull-up transistors and NMOS pull-down transistors. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a circuit diagram of a conventional CMOS inverter; 
     FIG. 2 is a circuit diagram of a 2-input NAND gate having PMOS and NMOS transistors sized in a conventional manner; 
     FIG. 3 is a block diagram of a dynamic flip-flop driving static logic gates sized in accordance with one embodiment of the present invention; 
     FIG. 4 is a timing diagram illustrating the pre-charge and evaluation phases of the dynamic flip-flop of FIG. 3; 
     FIG. 5 is a block diagram of a plurality of dynamic flip-flops driving a multiple-input embodiment of the static logic circuit of FIG. 3; 
     FIG. 6 is a circuit diagram of a 2-input NAND gate having PMOS and NMOS transistors sized in accordance with one embodiment of the present invention; 
     FIG. 7 is a circuit diagram of an N-input NAND gate having PMOS and NMOS transistors sized in accordance with another embodiment of the present invention; and 
     FIG. 8 is a circuit diagram of an N-input NOR gate having PMOS and NMOS transistors sized in accordance with yet another embodiment of the present invention. 
     Like reference numerals refer to corresponding parts throughout the drawing figures. 
    
    
     DETAILED DESCRIPTION 
     Embodiments of the present invention are discussed below in the context of static logic being driven by dynamic flip-flops for simplicity only. It is to be understood that embodiments of the present invention are useful in re-sizing PMOS pull-up devices for any application having a number of input signals that simultaneously transition to the same logic state. Accordingly, the present invention is not to be construed as limited to specific examples described herein but rather includes within its scope all embodiments defined by the appended claims. 
     FIG. 3 is a block diagram of a system  30  including a dynamic flip-flop  31  and a static logic circuit  32  having transistors sized in accordance with the present invention. The dynamic flip-flop  31  is well-known and may be, for example, of the type disclosed in U.S. Pat. No. 5,825,224 issued to Klass et al on Oct. 20, 1998 and assigned to the assignee of the present invention, incorporated by reference herein, although other dynamic flip-flops may be used. The dynamic flip-flop  31  has an input terminal coupled to receive an input signal X, and output terminals to provide complementary output signals x and {overscore (x)} to the logic circuit  32 . A clock signal CLK provided to respective clock input terminals of the dynamic flip-flop  31  and logic circuit  32  defines alternating pre-charge and evaluation phases for the system  30 . During the pre-charge phase, the dynamic flip-flop  31  drives output signals x and {overscore (x)} to a first logic state, e.g., logic low, and during the evaluation phase, the dynamic flip-flop  31  samples the input signal X and, in response thereto, drives either output signal x or its complement {overscore (x)} to a second logic state, e.g., logic high, as illustrated, for instance, in FIG.  4 . 
     Although shown in FIG. 3 as receiving a single input signal x and its complement {overscore (x)}, in actual embodiments the logic circuit  32  may receive a plurality of input signals (and/or corresponding complementary input signals), whereby each input signal is driven by a corresponding dynamic flip-flop  31  to ensure compatibility with dynamic logic, e.g., to ensure that the input signals are driven to the logic low state during each pre-charge phase. For example, FIG. 5 shows a plurality of dynamic flip-flops  31  connected between a respective plurality of input signals X( 0 )-X(n) and the logic circuit  32 , where each dynamic flip-flop  31  receives a corresponding one of the input signals X( 0 )-X(n) and, in response thereto, drives respective complementary signals x( 0 )-x(n) and {overscore (x)}( 0 )-{overscore (x)}(n) in the manner described above with respect to FIG.  4 . 
     Logic circuit  32  includes a number of suitable logic gates such as, for instance, AND gates, NAND gates, OR gates, NOR gates, XOR gates, XNOR gates, and so on. Since the input signals to the logic circuit  32  are driven to logic low during each pre-charge phase, the PMOS pull-up devices that form charge paths for gates within the logic circuit  32  may be sized in accordance with the present invention. Specifically, since all input signals simultaneously transition to logic low during each pre-charge phase, all PMOS pull-up devices simultaneously turn on to charge their respective output nodes toward a supply voltage during each pre-charge phase. Thus, each PMOS pull-up device connected in parallel between the supply voltage and the output node always participates in charging the output node toward the supply voltage. As a result, the effective drive strength for the charge path is equal to the sum of the individual drive strengths of all the PMOS pull-up devices, rather than equal to the drive strength of just one of the pull-up devices. This allows the size of each such PMOS pull-up device to be reduced while maintaining equal drive strengths between the PMOS charge path and NMOS discharge path. 
     For example, FIG. 6 shows a 2-input NAND gate  60  that is one embodiment of the logic circuit  32  of FIG. 5 having PMOS pull-up and NMOS pull-down devices sized in accordance with the present invention. The NAND gate  60  includes PMOS pull-up transistors  61  and  62  connected in parallel between the output node  63  and the supply voltage V DD , and NMOS pull-down transistors  64  and  65  connected in series between output node  63  and ground potential. The input signals x 0  and x 1  are dynamic signals provided by corresponding dynamic flip-flops (not shown in FIG. 5 for simplicity), and are thus driven to logic low during the pre-charge phase of the clock cycle. NAND gate  60  is a static logic circuit, and does not receive the complementary input signals {overscore (x 0 )}-{overscore (xn)} shown in FIG.  5 . 
     Since the NMOS transistors  64  and  65  are connected in series between output node  63  and ground potential, the total resistance of the discharge path is approximately twice the resistance of each NMOS transistor  64  and  65 . Accordingly, the effective drive strength S n  of the NMOS pull-down path is approximately one-half that of a single NMOS transistor, i.e., S n ≅W n /2. However, unlike the prior art NAND gate  20  of FIG. 2, the PMOS pull-up transistors  61  and  62  of the NAND gate  60  simultaneously turn on during each pre-charge phase since the dynamic input signals x 0  and x 1  are driven to the logic low state during each pre-charge phase. Hence, the effective drive strength for the charge path is the sum of the drive strengths of the PMOS transistors  61  and  62 , i.e., S p ≅2W p . Therefore, in order to maintain equal drive strengths for the charge and discharge paths, where S p ≅2W p =S n ≅W n /2, the width of each PMOS transistor  61  and  62  is given by W p =βW n /4. Accordingly, the present invention allows the size of PMOS pull-up transistors  61  and  62  of the logic gate  60  to be approximately one-half that of the PMOS pull-up transistors of the prior art NAND gate  20  of FIG. 2, for which the PMOS pull-up transistors  21  and  22  are sized as W p =βW n /2. 
     The sizing of parallel-connected PMOS pull-up devices according to present embodiments may be more generally expressed as W p =βW n /Nk, where N is the number of NMOS transistors connected in series between the output node and ground potential, and k is the number of PMOS pull-up devices that simultaneously turn on to charge the output node. For example, FIG. 7 shows an N-input NAND gate  70  having N PMOS pull-up transistors  71 ( 0 )- 71 (N−1) connected in parallel between the output node  72  and the supply voltage V DD , and N NMOS pull-down transistors  73 ( 0 )- 73 (N−1) connected in series between the output node  72  and ground potential. The PMOS pull-up transistors  71 ( 0 )- 71 (N−1) and NMOS pull-down transistors  73 ( 0 ) 73 (N−1) have respective gates coupled to receive corresponding input signals x( 0 )-x(N−1). The effective drive strength of the discharge path to ground potential through the N NMOS transistors  73 ( 0 )- 73 (N−1) is inversely proportional to the number of series-connected NMOS pull-down transistors between output node  72  and ground potential, i.e., S n ≅W n /N. The effective drive strength of the charge path to V DD  via the parallel-connected PMOS transistors is proportional to the number of PMOS pull-up transistors  71 ( 0 )- 71 (N−1) which turn on simultaneously, i.e., S p ≅kW p . Thus, sizing the PMOS pull-up transistors to a width of W p =βW n /Nk facilitates equal drive strengths for the charge and discharge paths. Note here that k may be any integer greater than 1 and less than or equal to N, and therefore the present invention is also applicable where some but not all of the input signals simultaneously transition to the same logic state. 
     Sizing schemes in accordance with the present invention are in contrast to conventional sizing schemes which size the PMOS pull-up transistors as W p =βW n /N. Accordingly, sizing PMOS pull-up transistors in accordance with the present invention allows for PMOS pull-up transistors to be reduced by the factor k while preserving equal drive strengths for the charge and discharge paths. Thus, for instance, PMOS pull-up device width accordance with the present invention by a factor of 4 over prior art NAND gates. 
     The present invention is also applicable to logic circuits which have a number of NMOS pull-down transistors connected in parallel between the output node and ground potential and a number of PMOS pull-up devices connected in series between the output node and the supply voltage V DD . For example, FIG. 8 shows an N-input NOR gate  80  having N PMOS pull-up transistors  81 ( 0 )- 81 (N−1) connected in series between an output node  82  and V DD  and N NMOS pull-down transistors  83 ( 0 )- 83 (N−1) connected in parallel between output node  82  and ground potential. The PMOS pull-up transistors  81 ( 0 )- 81 (N−1) and NMOS pull-down transistors  83 ( 0 )- 83 (N−1) have respective gates coupled to receive corresponding input signals x( 0 )-x(N−1), as shown in FIG.  8 . The effective drive strength of the charge path to V DD  through the N PMOS transistors  81 ( 0 )- 81 (N−1) is inversely proportional to the number of series-connected PMOS pull-up transistors between output node  82  and V DD , i.e., S p ≅W p /N. The effective drive strength of the discharge path to ground potential via the parallel-connected NMOS transistors is proportional to the number of NMOS pull-down transistors  83 ( 0 )- 83 (N) which turn on simultaneously, i.e., S n ≅kW n . Thus, sizing the PMOS pull-up transistors to a width of W p =NβW n /k facilitates equal drive strengths for the charge and discharge paths. 
     The down-sizing of PMOS pull-up devices facilitated by the present invention may result in significant savings in silicon area and power consumption, especially in applications having large arrays of logic gates that are driven by dynamic flip-flops or by any other circuit that simultaneously switches a number of input signals to the same logic state. For example, since row decoders used to decode addresses for memory devices such as SRAM and register files typically include hundreds or even thousands of 2-input and/or 3-input logic gates, reducing the size of PMOS pull-up devices used in such row decoders by respective factors of 2 and/or 3 as described above in accordance with the present invention can significantly reduce the silicon area occupied by such row decoders. 
     While particular embodiments of the present invention have been shown and described, it will be obvious to those skilled in the art that changes and modifications may be made without departing from this invention in its broader aspects and, therefore, the appended claims are to encompass within their scope all such changes and modifications as fall within the true spirit and scope of this invention.