Abstract:
A radio frequency voltage controlled oscillator and method for designing it are provided. The RF VCO comprises a differential oscillator and a cascoded current source. The cascoded current source substantially provides a constant current bias to the differential oscillator. A first biased transistor in the cascoded current source is connected to the differential oscillator. A second biased transistor is cascoded to the first biased transistor. A low pass filter is cascoded between the first second biased transistors.

Description:
BACKGROUND OF THE INVENTION 
       [0001]    1. Field of the Invention 
         [0002]    The invention generally relates to radio frequency (RF) voltage controlled oscillators (VCO) and in particular to VCOs with reduced phase noise. 
         [0003]    2. Description of the Related Art 
         [0004]    RF communications such as cellular phone applications rely on analog circuits to generate various channel frequencies. Thus, voltage controlled oscillators (VCO) have become important elements of RF communication devices, such as transmitters, where VCOs are used as master oscillators, and receivers, where VCOs are used as local oscillators. 
         [0005]    One of the major obstacles to full integration of VCOs is the high phase noise level generated when VCOs are embedded in a frequency synthesizer. Phase noise is rapid, short-term, random fluctuation in the phase of a wave, caused by time domain instabilities and found mostly in active elements used in VCOs. This low frequency noise signal source is often referred to as flicker noise, or noise, in bipolar and Metal Oxide Semiconductor (MOS) transistors. 
         [0006]    In comparison with those biased by constant voltage, VCOs biased by constant current are superior to supply or ground common-mode fluctuation, insensitive to process corner variation, and capable of higher output voltage swing. The noise in the active devices of current sources, nevertheless, can be up-converted into an LC tank of a VCO, thereby aggravating the phase noise. To prevent the up-conversion of the noise, various active and/or passive devices are used. 
         [0007]      FIG. 1  is a circuit diagram of a conventional RF VCO, in which a biased bipolar transistor Bd is employed as a constant tail current source to bias a differential oscillator. Bipolar transistors are known to have little noise. Even though the noise of bipolar transistor Bd in  FIG. 1  is up-converted into the differential oscillator, its effects on the phase noise of the VCO may be negligible. The manufacturing of bipolar transistors together with CMOS transistors is, however, generally more complex and costly than that of only CMOS transistors. 
         [0008]      FIG. 2  is a circuit diagram of another conventional RF VCO. Referring to  FIG. 2 , a low pass filter, including an inductor Ld and a capacitor Cd, is interposed between a differential oscillator and a current source, such that the low pass filter filters the noise from the drain of the current source. The inductor Ld or the capacitor Cd must be large in order to effectively drain the noise to ground, so the implementation of the inductor or capacitor is impractical. 
       BRIEF SUMMARY 
       [0009]    A radio frequency voltage controlled oscillator is provided, comprising a differential oscillator and a cascoded current source. The cascoded current source substantially provides a constant current bias to the differential oscillator. A first biased transistor in the cascoded current source is connected to the differential oscillator. A second biased transistor is cascoded to the first biased transistor. A low pass filter is cascoded between the first second biased transistors. 
         [0010]    A method of designing a radio frequency voltage controlled oscillator is provided. A differential oscillator and a cascoded current source are arranged, such that the cascoded current source substantially provides a current bias required to drive the differential oscillator. The cascoded current source comprises two biased active devices and a low pass filter. One of the two biased active devices is cascoded to the other. The low pass filter is connected between the two biased active devices. 
         [0011]    A detailed description is given in the following embodiments with reference to the accompanying drawings. 
     
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         [0012]    The invention can be more fully understood by reading the subsequent detailed description and examples with references made to the accompanying drawings, wherein: 
           [0013]      FIGS. 1 and 2  are circuit diagrams of two conventional RF VCOs, 
           [0014]      FIG. 3  is a circuit diagram of a RF VCO; 
           [0015]      FIG. 4  is a circuit diagram of a RF VCO according to embodiments of the invention; 
           [0016]      FIG. 5  is another circuit diagram of a RF VCO; 
           [0017]      FIG. 6  illustrates phase noise of the output signal versus the frequency offset from the fundamental frequency f 0  of RF VCOs in  FIGS. 3-5 , respectively; and 
           [0018]      FIG. 7  shows output waves  72 ,  74  and  76  probed from the outputs of RF VCOs in  FIGS. 3-5 ; and 
           [0019]      FIGS. 8 and 9  are circuit diagrams of other RF VCOs according to embodiments of the invention. 
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0020]    The following description is of the best-contemplated mode of carrying out the invention. This description is made for the purpose of illustrating the general principles of the invention and should not be taken in a limiting sense. The scope of the invention is best determined by reference to the appended claims. 
         [0021]      FIG. 3  is a circuit diagram of a RF VCO  10 , comprising a differential oscillator  12  and a cascoded tail current source  14 . Inductors L 1  and L 2 , and capacitors C 1  and C 2  form a LC tank  16 , substantially determining the fundamental resonant frequency f 0  of RF VCO  10 . The connecting node between capacitors C 1  and C 2  acts as a frequency control terminal of RF VCO  10 , voltage on which changes the capacitances of capacitors C 1  and C 2 , thereby determining the fundamental resonant frequency f 0 . A cross-coupled transistors pair  18 , including MOS transistors M 1  and M 2 , forms a feedback circuit of RF VCO  10 , such that the gate of MOS transistor M 1  is connected to the drain of MOS transistor M 2 , and the gate of MOS transistor M 2  to the drain of MOS transistor M 1 . Cascoded tail current source  14  consists of MOS transistors MS 1  and MS 2 , substantially providing a constant current to drive differential oscillator  12 . 
         [0022]    It is preferred that MOS transistor MS 2  have longer and wider channel and MOS transistor MS 1  shorter and narrower. Thus, as noise is in positive relationship with current density through a MOS channel, MOS transistor MS 2  has relatively insignificant noise. MOS transistor MS 1  has more significant noise, which nevertheless will be rejected or alleviated by the cascoded configuration and causes little phase noise to the output wave from differential oscillator  12 . Furthermore, less channel length and width also form a small parasitic drain capacitor, lessening the capacitive loading of cascoded tail current source  14  and making it a more ideal current source, with no capacitive loading. 
         [0023]    The impedance of MOS transistor MS 2  significantly drops at a higher frequency, more particularly due to its channel length and width which form a large parasitic capacitor connected to an ac ground. This large parasitic capacitor effectively connects or shorts the source of MOS transistor MS 1  to the ac ground at a higher frequency, and the cascoded configuration in  FIG. 3  appears to malfunction since MOS transistor MS 2  provides very little effective impedance and cannot boost the overall output impedance seen from differential oscillator  12 . Absence of cascoded configuration implies no rejection of significant noise from MOS transistor MS 1 . In other words, the phase noise of RF VCO  10  at a higher frequency is aggravated. 
         [0024]      FIG. 4  is a circuit diagram of another RF VCO  20 , comprising a differential oscillator  12  and a cascoded tail current source  24 .  FIG. 4  differs from  FIG. 3  only in the presence of an additional inductor LS as a low pass filter connected between cascoded MOS transistors MS 1  and MS 2 . To differentiate the cascoded tail current  14  in  FIG. 3 , the cascoded tail current source in  FIG. 4  is denoted by number  24 . At a lower frequency, the impedance of inductor LS, equal to jωL, is negligible, effectively shorting the source of MOS MS 1  to the drain of MOS MS 2 . Thus, the circuit configuration in  FIG. 4  is the same as that in  FIG. 3  at a lower frequency. Inductor LS plays a key role at a higher frequency, boosting the overall output impedance of cascoded tail current  24 . Even though the large parasitic capacitor in MOS transistor MS 2  effectively shorts the drain of MOS transistor MS 2  to an ac ground at a higher frequency as mentioned, inductor LS, with impedance raises as frequency increases, still significantly stands between the source of MOS transistor MS 1  and the ac ground. The overall output impedance of cascoded tail current source  24  is g m r 0 (jωL), where g m  and r 0  are properties inherent to MOS transistor MS 1  and L is the inductance of inductor LS. Irrespective of the condition at a higher or lower frequency, the noise from MOS transistor MS 1  is rejected or alleviated by the cascoded configuration effectively contributed by inductor LS or MOS transistor MS 2 . In conclusion, RF VCO  20  in  FIG. 4  is more immune to the noise from MOS transistor MS 1  than RF VCO  10  in  FIG. 3 . 
         [0025]      FIG. 5  is a circuit diagram of a voltage-biased RF VCO  30 . Unlike the current-biased RF VCOs  10  and  20  in  FIGS. 3 and 4 , RF VCO  30  in  FIG. 5  lacks a tail current source and has only differential oscillator  12  directly powered by power rails Vdd and ground. 
         [0026]      FIG. 6  plots  62 ,  64  and  66 , illustrating the phase noise of the output signal versus the frequency offset from the fundamental frequency f 0  of RF VCOs  10 ,  20  and  30 , respectively. As can be seen, the phase noises of current-biased RF VCOs  10  and  20  at 10 KHz offset frequency (a lower offset frequency) are substantially the same, but higher than that of voltage-biased RF VCO  30  because of the noise of MOS transistor MS 2  in current-biased RF VCOs  10  and  20 . At a higher offset frequency, such as 20 MHz, plots  62  and  64  separate, and the phase noise of current-biased RF VCO  20  is improved about 3 dB compared with that of current-biased RF VCO  10 , which lacks inductor LS. 
         [0027]      FIG. 7  shows output waves  72 ,  74  and  76  probed from the outputs of RF VCOs  10 ,  20  and  30 . As can be seen, wave  74 , corresponding to current-biased RF VCO  20  with inductor LS, has an output voltage swing of about 2.8, the largest of the 3 waves  72 ,  74  and  76  in  FIG. 7 . 
         [0028]    As shown in  FIG. 4 , it is critical for inductor LS to be located between MOS transistors MS 1  and MS 2 . Otherwise, locating an inductor between the source of MOS transistor MS 2  and ground barely affects the output impedance of the cascoded tail current source at a higher frequency because of the large grounded parasitic capacitor of intervening MOS transistor MS 2 , which effectively shorts one terminal of the inductor to ground. An inductor between the drain of MOS transistor MS 1  and differential oscillator  12  does not experience the output impedance boost caused by the gain stage of MOS transistor MS 1  at a higher frequency. To have the same impedance as that between MOS transistors MS 1  and MS 2  at a higher frequency, an inductor connected to the drain of MOS transistor MS 1  requires much higher inductance, occupying more silicon surface. Thus, it is preferred, economically and practically, for inductor LS to be connected in serial between MOS transistors MS 1  and MS 2 . 
         [0029]    Each of both cascoded current sources  14  and  24  in  FIGS. 3 and 4  is connected between ground and a differential oscillator. The invention is not limited thereto, however. Embodiments of the invention may have a current source connected between VDD power line and a differential oscillator, as shown in  FIG. 8  where cascoded current source  26  has cascoded MOS transistors MD 1  and MD 2 , and an inductor LD therebetween. It is preferred that MOS transistor MD 2  has a longer and wider channel than MOS transistor MD 1 . 
         [0030]    A differential oscillator in an embodiment of the invention may be different from those disclosed in  FIGS. 1-5  and  8 .  FIG. 9  exemplifies another RF VCO  40  with an alternative differential oscillator, which has a cross-coupled transistor pair (including MOS transistors M 3  and M 4 ) connected to inductors L 1  and L 2  and VDD power line. Each of capacitors C 1  and C 2  may comprise a varactor with voltage-controllable capacitance for frequency tuning. 
         [0031]    As the phase noise of RF VCO  20  is insensitive to the inductance variation of inductor LS, it is unimportant to have an inductor with a highly-accurate inductance, such that a multi-turn or 3D inductor is acceptable. Furthermore, the metal line width used in inductor LS can be smaller since substantially constant current flows therethrough. 
         [0032]    While the invention has been described by way of examples and in terms of preferred embodiment, it is to be understood that the invention is not limited thereto. To the contrary, it is intended to cover various modifications and similar arrangements (as would be apparent to those skilled in the art). Thus, the scope of the appended claims should be accorded the broadest interpretation so as to encompass all such modifications and similar arrangements.