Abstract:
A signal isolator using magnetic coupling is disclosed. In contrast to the prior art non-isolated voltage summing circuits, the present invention utilizes current summing and magnetic coupling. In addition to providing ground and signal isolation the circuit of the present invention also provides a current-summing node which is always in a high impedance state, thereby allowing ancillary control mechanisms to be easily implemented in the circuit.

Description:
FIELD OF THE INVENTION 
     This invention relates to the field of signal ground isolation and, more particularly, to signal ground isolation using magnetic coupling. 
     BACKGROUND OF THE INVENTION 
     In order to provide electrical isolation, e.g., for safety considerations, most switching DC-DC power converters employ conventional optocouplers. FIG. 1 illustrates a prior art optocoupler circuit  100 . A steady DC voltage V IN  received from the output of a main switching converter (not shown) is scaled by a resistor network  101  and  103  and is compared to a reference voltage V REF  via a high gain amplifier  107 . Grounding point  102  comprises the power return path. The high gain amplifier  107  compares the scaled V IN  and V REF  and outputs an error signal voltage V ERR , representing the difference between V IN  and V REF . The error signal voltage V ERR  drives an LED  111 , causing LED  111  to emit light across isolation barrier  113  to a photo-transistor  119 . Grounding point  122  comprises the power return path for this side of the circuit. In a known manner, photo-transistor  119  converts the light emitted from LED  111  back to a current signal representing the difference between the scaled V IN  and V REF , which is converted by resistor  117  back to a voltage representing the error signal voltage V ERR . Resister  105  converts the bias current input to or output from high gain amplifier  107 , thereby balancing any input bias-voltage imbalance. 
     The optocoupler circuit  100  of FIG. 1 provides isolation across isolation barrier  113 ; however, it operates with a restricted temperature range because the semi-conductor junction materials of photo-transistor  119  can only withstand temperatures between −20° centigrade and 95° centigrade, thereby limiting the dynamic range of the circuit. In addition, since the light emitted by light emitting diodes such as LED  111  is relatively weak in intensity, the isolation barrier  113  between LED  111  and photo-transistor  119  must be kept relatively small. Due to the close proximity of LED  111  with respect to photo-transistor  119 , capacitive coupling can occur between the two devices, thereby introducing AC coupling between the two devices and degrading the isolation that they provide. 
     It is also well known to utilize transformers to provide isolation between two electrical circuits so as to isolate a source of relatively high voltage that powers a device from low voltage devices and/or from a user of the isolated device. For example, isolation transformer are commonly used in medical equipment, such as temperature monitors, electro-cardiograms, oximeters, or invasive blood pressure monitors which include sensors which are in contact with the patient. U.S. Pat. No. 5,615,091, for example, incorporated fully herein by reference, is directed to an isolation transformer for medical equipment. 
     Non-isolated current sampling voltage summing circuits are also known. FIG. 2 illustrates a prior art current sampling voltage summing circuit  200 . As shown in FIG. 2, a resistor-divider comprising resistors  201  and  203  scales a steady DC voltage V IN  from a main switching converter and provides an input to a comparator, e.g. high gain amplifier  207 . Grounding points  202  comprise the power return path for this side of the circuit. High gain amplifier  207  compares this input with a reference voltage V REF  input via resistor  205 , just as in FIG.  1 . The output V ERR  of high gain amplifier  207  is an error voltage signal which is applied to the base of transistor  223  via resistor  221 . Transistor  223  acts as a voltage follower, since the error signal voltage V ERR  will go across the base-emitter junction of transistor  223  and “sit” on top of the emitter. Thus, the emitter voltage of transistor  223  is the sum of the base-emitter junction voltage of the transistor  223  plus the error voltage V ERR , in volts. 
     A power transformer T 1  having a primary winding  227  and a secondary winding  229  is switchable between an energized and a de-energized state by switching transistor  225 . Grounding point  222  comprises the power return path for this side of the circuit. In a known manner, a non-isolated current sensing/sampling block  240  yields a current output kI P  that is a scaled version of primary current I P  at current input node  228  with the scaling factor k. The output current, kI P  passes an emitter resistor  209  and produces a pulsating voltage V SENSE . This pulsating voltage V SENSE  is added to the error voltage sitting at the emitter of transistor  223 , and the sum of these two voltages presents itself as a non-isolated feedback signal used in a pulse width modulator. By comparing the non-isolated feedback signal with another known reference voltage, a driving pulse with variable time duration (width) is provided for switching transistor  225 . However, due to the non-isolated nature of current sensing/sampling block  240 , grounding point  202  and grounding point  222  are in essence the same. 
     The magnetic coupling circuit  200  of FIG. 2 has certain deficiencies. It does not provide isolation and it must “overcome” the base-emitter junction voltage of transistor  223 ; thus it is unable to handle low-level signals. For example, if the primary current I P  is relatively small and the sampled current output kI P  multiplied by the ohmic value of the sensing resistor  209  is not large enough to overcome the base-emitter junction of transistor  223 , the circuit will not function because the circuit is, in effect, an open circuit. This will cause the control loop to be opened rendering it unable to control the converter output V IN . To properly function, the circuit must function at all times. 
     SUMMARY OF THE INVENTION 
     An object of the present invention is to provide a signal isolator using magnetic coupling. In contrast to the prior art non-isolated voltage summing circuits, the present invention utilizes current summing and magnetic coupling. In addition to providing ground and signal isolation the circuit of the present invention also provides a current-summing node which is always in a high impedance state, thereby allowing ancillary control mechanisms to be easily implemented in the circuit. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 illustrates a prior art optocoupler used as a signal isolator; 
     FIG. 2 illustrates a prior art employing non-isolated current sampling and voltage summation; 
     FIG. 3 illustrates an embodiment of the presentation comprising a magnetic isolator using current and current summation; 
     FIG. 4 illustrates an alternative embodiment of a portion of the circuit of FIG. 3; and 
     FIG. 5 illustrates an alternative embodiment of the circuit of FIG.  3 . 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     FIG. 3 illustrates a preferred embodiment of the circuit of the present invention. Referring to FIG. 3, a current summing circuit  300  includes a current summing block  340  and a reset block  350 . Grounding points  332  comprises the power return path for both blocks. Current summing block  340  is connected to the output of high gain amplifier  307 . In contrast to the prior art circuit of FIG. 2 which sums voltages, the circuit of FIG. 3 sums currents and also isolates the input V IN  of the circuit from the output V OUT . The current-summing centers around a specific point, high impedance node  360  (discussed in more detail below). As a result many different signal inputs can be presented to the circuit at node  360  without requiring reconfiguration of the circuit. By contrast, voltage summing, as exemplified by the prior art circuit of FIG. 2, involves a complete voltage summing loop (as opposed to a specific point). To enable a loop to accommodate different input signals, the loop must be broken and reconfigured. A one-turn winding  331  is fed through the toroid  333  of a current transformer T 2  and acts as the primary winding of current transformer T 2 . Current transformer T 2  has two secondary windings  335  and  337 . Thus, current in the primary of power transformer T 1  is also reflected to the secondary windings  335  and  337  of current transformer T 2  and, thus, current transformer T 2  operates as a current sensing transformer. Secondary winding  337  of current transformer T 2  senses a fraction of the primary current (called the “reflected current”) from power transformer T 1 , i.e., the primary current I P  of power transformer T 1  divided by the number of turns n of secondary winding  337  of current transformer T 2 . 
     The error voltage V ERR  output from high gain amplifier  307  is converted to a current I ERR  via transistor  323 . This current I ERR  is then summed with the reflected current from secondary winding  337  and “injected” across sensing resistor  326 . Thus, the voltage across sensing resistor  326  (the current-sum times the ohmic value of resistor  326 ) converts the summed currents into voltage, and secondary winding  335  acts as a voltage sampling winding. In essence, the pulsating voltage across secondary winding  335  carries information pertaining to both the voltage loop and the current loop. 
     Reset circuit  350  is required because when power transformer T 1  is energized (i.e. when power transformer T 1  has current I P  actively flowing) secondary windings  335  and  337  of current transformer T 2  will sense current. However, the primary side current of primary winding  327  alternates on and off due to the switching action of transistor  325 . Thus, when the current is off there is no current on the primary winding  327  of power transformer T 1  and, under this condition, it is necessary to reset current transformer T 2  so that it is ready for the next cycle to begin. Without resetting current transformer T 2 , the transformer core of current transformer T 2  will be saturated and current transformation will cease, opening the control loop. 
     When the power transistor  325  is turned off, the dotted ends of secondary windings  335  and  337  will change polarity, i.e., they will become negative in terms of voltage. When the dotted end of secondary winding  335  changes polarity and becomes negative, resistor  352  will have a negative voltage applied to it and will forward bias transistor  356 . When transistor  356  is forward biased, current flows through the transistor and this current flow resets the core  333  of current transformer T 2 ; at the same time the emitter current also discharges capacitor  346  which is situated in parallel with resistor  344 . Resetting the current summing block  340  is important because the capacitor  346  should always start from zero for every operating cycle since a non-zero starting voltage will upset the pulse-width modulation. 
     The circuit of FIG. 3 makes it possible to implement many additional features which cannot be implemented with the circuit of FIG.  2 . The current summing circuit  300  of FIG. 3 includes high impedance node  360 . High impedance node  360  is a current source point that is always at a high impedance; thus, a soft-start circuit  362  as shown in FIG. 3 can be connected to high impedance node  360  to, in a known manner, allow for a gradual power up of the circuit  300  and the entire converter. Similarly, an overload circuit  364  can be connected to the high impedance node  360  and, in a known manner, provide an output to shut the system down when an overload condition (e.g., the load current exceeding a specific limit) is reached. Many circuits providing current-form signals can be connected to high impedance node  360 ; the novelty lies in the provision of the high impedance node  360  and not in the soft-start circuit  362  or the overload circuit  364  themselves. 
     By way of example only, the following is a list of specific components that may be utilized to construct the circuit of FIG.  3 . 
     
       
         
               
               
             
           
               
                   
               
               
                 FIG. 3 ITEM 
                 PART NO. 
               
               
                   
               
             
             
               
                 Transistors 323, 356 
                 2N2907 (Motorola) 
               
               
                 Transformer 333 
                 F-41005-TC (Magnetics, Inc., Butler, PA) 
               
               
                 Amplifier 307 
                 LM158 (National Semiconductor) 
               
               
                 Diode 339 
                 1N6642 
               
               
                 Diode 324 
                 1N6640 
               
               
                 Secondaries 335, 337 
                 Magnetic Wire AWG#30, 50 turns, Bifilar 
               
               
                 All resistors other than 
                 RNC55XXXXFS, 1% 
               
               
                 resistor 326 
               
               
                 Resistor 326 
                 RNC55XXXXBS, 0.1% 
               
               
                   
               
             
          
         
       
     
     FIG. 4 illustrates an alternative structure for a portion of the current summing circuit of FIG. 3 . In FIG. 4, instead of using the PNP transistor  323  of FIG.3, an NPN transistor  423  is used. In addition, diode  323  is removed from the circuit. Because of this configuration, resistor  426  can be larger than resistor  326  of FIG.  3 . This circuit, instead of being a current summing current, is a current subtraction circuit. Because it is a current subtraction circuit, the dynamic range of the operating voltage is wider and, given the expected peak sensing current, the resistor  426  can be larger, the exact size depending on the overall design of the circuit. The use of a larger resistor results in less gain and therefore better loop stability. While this approach requires the use of a negative power supply, in certain applications (e.g., a power converter with a negative output) this negative control signal technique may be preferred. 
     FIG. 5 illustrates an alternative embodiment in which the single turn primary  331  of current transformer T 2  of FIG. 3 is moved to the secondary side of power transformer T 1 . By placing the single turn primary  331  of current transformer T 2  on the secondary side of the transformer T 1 , the same functions of the FIG. 3 circuit are preserved, but with a different scale factor since the secondary current of T 1  is proportional to the turn ratio of T 1 . 
     In addition to the many benefits described above, the present invention also provides advantages related to loop stability. As mentioned above, the current-transfer-ratio (CTR) of an optical isolator circuit is susceptable to temperature drift. Thermal drift of the CTR eventually results in loop gain fluctuation and consequently, loop instability. Since the current-summing magnetic isolator of the present invention offers a stable coupling coefficient, this also eliminates temperature-induced loop instability. 
     In addition, conventionally current-mode control schemes compare a pulsating current signal with the optically isolated voltage-loop feedback. Due to the limited filtering capability, the optical isolated voltage-loop feedback carries a low amplitude AC component. Due to the AC component, under some transient conditions (e.g. initial start-up or sudden input or load changes) subharmonic modulation can lock the loop in a sub harmonic modulation mode. This effect will present itself as alternating long and short duty cycles, which results in a frequency at approximately one-half the normal clock cycle. This half-frequency (subharmonic) results in a larger output ripple voltage because the output filter may not be designed to attenuate this voltage component. In accordance with the present invention, instead of comparing the voltage feedback with a pulsating current signal, the present invention compares the composite feedback with a constant dc reference voltage. This eliminates the possibility of the circuit becoming locked into an undesirable operation mode such as a subharmonic modulation mode. 
     While there has been described herein the principles of the invention, it is to be understood by those skilled in the art that this description is made only by way of example and not as a limitation to the scope of the invention. Accordingly, it is intended by the appended claims, to cover all modifications of the invention which fall within the true spirit and scope of the invention.