Abstract:
A vehicle traffic sensor for detecting and monitoring vehicular targets is presented. The sensor employs a planar design resulting in a reduced profile sensor. The sensor includes a multi-layer radio frequency board with RF components on one of the sides and both isolation and planar array antennas on the opposing side. The antennas are preferably tapered planar array antennas which include one transmit antenna and one receive antenna. The sensor also includes at least one logic or signal processing board populated with components on a first side and a ground plane on a second side positioned toward the RF componentry of the RF board to form an RF shield. The boards are housed within a housing that is permeable, at least on the side through which the antenna structures propagate.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS  
       [0001]    The present application is a continuation of U.S. patent application Ser. No. 09/964,668, filed on Sep. 27, 2001, which is incorporated herein by reference. 
     
    
     
       BACKGROUND OF THE INVENTION  
         [0002]    1. The Field of the Invention  
           [0003]    The present invention relates generally to vehicular traffic monitoring systems, and more particularly relates to sensors for detecting the presence, location, speed, direction of travel, volume, and occupancy of vehicular traffic on a roadway.  
           [0004]    2. The Relevant Technology  
           [0005]    Controlled signalized intersections represent a key element in urban planning, public safety and traffic control. The science and engineering of traffic planning and control has long relied on the use of sensor devices designed for this specific purpose and, more recently, for the collection of traffic flow data. Some of these device technologies, such as those embedded in the roadways, have been employed for over sixty years and continue to require the same amount of attention in installation, calibration, maintenance, repair and replacement as they did decades ago. This laborious caretaking can be due to a number of factors ranging from inferior product design and poor installation to post installation disruption and migratory changes in traffic flow patterns. Reliability of these technologies is an issue to an overall traffic control plan and can prove extremely costly to maintain as an integral component to an overall traffic plan.  
           [0006]    Traffic control devices that are embedded in roadways serve the interest of public safety, but in the event of a new installation, or maintenance/repair, they act as a public nuisance, as repair crews are required to constrict or close multiple lanes of traffic for several hours to reconfigure a device, or even worse, dig up the failed devices for replacement causing closure of the lane for several days or weeks.  
           [0007]    While several sensor technologies are employed to assist in traffic planning and control, the oldest and most widely used technology currently employed in controlled intersections is the inductive loop. This loop is an in-pavement fixed location sensor, with the limitation of sensing only the traffic that is immediately over it. While such devices have continued history of use, failures of loops are common and at any one time as many as 20%-30% of all installed controlled intersection loops are non-responsive. Furthermore, the cost to repair these devices can be greater than the original installation cost.  
           [0008]    As technology has developed over the decades, new sensory devices have been introduced to the traffic control industry. In recent years, there have emerged several non-intrusive technologies for traffic sensing that employ a remote sensor (i.e., not embedded in the roadway) as illustrated in FIG. 1. While the majority of these types of sensors  110  incorporate microwave radar technology, other types including optical devices have also taken hold. For example, intersection traffic cameras may be manually configured to analyze specific user-defined traffic zones at all times. As cameras rely on optics, (i.e., the ability to visually see the traffic that is to be monitored) they are susceptible to the forces of nature that can occlude visibility. These forces include sun glare, accumulated snow or dirt and darkness. Under ideal conditions cameras would only need to be serviced or reconfigured with major intersection redesign. Presently available systems require on-site attention to improve and upgrade the capability of the unit, or complete replacement for upgrading the camera itself.  
           [0009]    Another type of above-ground sensor includes acoustic sensors which operate as traffic sound-based listening devices. These devices employ an array of microphones built into the sensor allowing the device to detect traffic based on spatial processing changes in sound waves received at the sensor. After processing and analysis of the received sound waves, detection and traffic flow information is then assigned to the appropriate user-defined regions or lane being monitored forming a picture of the traffic.  
           [0010]    When acoustic sensors are deployed, their microphone sensitivity is pre-set for normal operating conditions which include typical weather conditions. Again, the software and operating instructions to control an acoustic sensor require on-site attention to improve and upgrade the capability of the unit, or complete replacement to upgrade the sensor itself.  
           [0011]    Other popular sensor types are based on microwave radar technology. Such sensors detect traffic based on the reflection of a transmitted electromagnetic signal depicted in FIG. 1 as signals  118 . The received signal is then processed into detection and traffic flow information which is then assigned to the appropriate user defined lane being monitored. As illustrated in FIGS. 2 and 3, microwave radar technology utilizes several bulky, expensive and manufacturably inefficient components to sense traffic. Most notably, microwave radar sensors are comprised of a mechanically-large horn antenna  170  and separate radio frequency components and controller boards that are individually tuned and matched in order to result in an operable system  180 . Furthermore, the unit requires on-site maintenance and attention to reconfigure, or upgrade software.  
           [0012]    As identified above, many useful forms of technology exist to monitor and detect traffic. However, many forms of detection are obtrusively bulky, manufacturing intense, and all require on site maintenance and attention to re-configure the software, or operating instructions when traffic conditions, climate, or other operating conditions change. Without reconfiguration, the devices will continue to sense, but with reduced accuracy and in the worst case they may discard the actual flow pattern as peripheral noise. The cost to manufacture and reconfigure devices can be costly, and disruption to traffic is common.  
         BRIEF SUMMARY OF THE INVENTION  
         [0013]    A vehicle sensor for detecting and monitoring vehicular targets is presented. The sensor employs a planar design resulting in a reduced profile sensor and a greatly improved sensor for manufacturing. Improvements are a result of controlled manufacturing processes for forming controlled interconnects and structures on replicable circuit boards.  
           [0014]    The sensor of the present invention includes a multi-layer radio frequency board having a first side which includes at least a majority of the RF components. On the opposing side of the board is a ground plane providing isolation to the RF components. Additionally, the opposing side also has printed thereon array transmit and receive antennas for radiating a signal toward a vehicular target and for receiving the signal as reflected from the vehicular target. The planar antennas provide a replicable antenna structure that is easily manufactured.  
           [0015]    The sensor device further includes logic/control functionality which may be colocated or positioned separately on at least one logic or signal processing board that is preferably populated with components on a first side with a ground plane on a second side. The second or ground plane side is preferably positioned toward the RF componentry of the RF board to form an RF shield about the RF componentry. The boards are housed within a housing that is permeable to electromagnetic waves, at least on the side through which the antenna structures radiate. To provide additional RF absorption and isolation, an RF absorber is placed between the boards to provide additional isolation of RF emanations near to the source of generation.  
           [0016]    These and other objects and features of the present invention will become more fully apparent from the following description and appended claims, or may be learned by the practice of the invention as set forth hereinafter.  
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0017]    To further clarify the above and other advantages and features of the present invention, a more particular description of the invention will be rendered by reference to specific embodiments thereof, which are illustrated, in the appended drawings. It is appreciated that these drawings depict only typical embodiments of the invention and are therefore not to be considered limiting of its scope. The invention will be described and explained with additional specificity and detail through the use of the accompanying drawings in which:  
         [0018]    [0018]FIG. 1 illustrates an above-ground sensor employing propagation delay calculation for position calculation of vehicular traffic;  
         [0019]    FIGS.  2 - 3  illustrate a radar sensor comprised of horn antennas and multiple modules, in accordance with the prior art;  
         [0020]    [0020]FIG. 4 illustrates an integrated above-ground traffic sensor, in accordance with the preferred embodiment of the present invention;  
         [0021]    [0021]FIG. 5 illustrates the mechanical integration of RF components and signal processing components, in accordance with the preferred embodiment of the present invention;  
         [0022]    [0022]FIG. 6 illustrates planar antennas integrated into the RF module board, in accordance with the preferred embodiment of the present invention;  
         [0023]    [0023]FIG. 7 is a detail of one of the planar antennas, in accordance with a preferred embodiment of the present invention;  
         [0024]    [0024]FIG. 8 is a block diagram of the component side of the RF board assembly and other related functional blocks, in accordance with the present invention;  
         [0025]    [0025]FIG. 9 is a detailed layout of the RF component side of the RF component side of the RF board, in accordance with the preferred embodiment of the present invention;  
         [0026]    [0026]FIG. 10 illustrates an embodiment of the present invention that employs a modulated signal digital generator for generating the desired signal;  
         [0027]    [0027]FIG. 11 illustrates an embodiment employing a mixer configuration for the digitally generated modulated signal generator, in accordance with an embodiment of the present invention; and  
         [0028]    [0028]FIG. 12 illustrates direct digital signal generation of the transmit signal, in accordance with another embodiment of the present invention.  
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0029]    [0029]FIG. 4 is a perspective view of a traffic monitoring sensor, in accordance with a preferred embodiment of the present invention. A sensor  500  is illustrated having a generally planar topology due to the planarization of components including planarization of a transmit and receive antenna. Sensor  500  is a generally sealed enclosure comprised of a material that is permissive to the exchange of electromagnetic propagations. Sensor  500  is also generally comprised of a housing  502  for enclosing the multi-layer radio frequency circuit board and other processing component boards such as digital signal processing and/or control assemblies. Housing  502  includes a back surface  504  and a top or front surface  506 . During deployment of sensor  500 , front surface  506  is directed generally orthogonal with the roadway or portion of roadway undergoing monitoring. In the present invention, the term “above-ground” sensor means that the sensor is not embedded into the roadway but rather may be mounted above or about the roadway at various acceptable angles.  
         [0030]    As the electromagnetic signals must propagate through front surface  506  as radiated from planar printed circuit board antennas described below, front surface  506  further includes geometries that facilitate reduced distortion of the antenna radiation pattern throughout the entire beamwidth of the antennas. FIG. 4 depicts such distortion-minimizing geometries as radomes  508  and  510 .  
         [0031]    [0031]FIG. 4 further depicts additional sensor enhancements which are structurally depicted as communication link antenna  512  which facilitates both transmission of accumulated sensor data as well as reception of commands and software upgrades. The power and serial communication link are facilitated through the connector  514 .  
         [0032]    [0032]FIG. 5 illustrates an exploded view of sensor  500 , in accordance with the preferred embodiment of the present invention. In addition to including housing  502 , sensor  500  further includes a multi-layer radio frequency circuit board  520  which includes radio frequency components  522  disposed on a first side  524 . Circuit board  520  further includes antennas  526  on an opposing second side  528 . Such a configuration accommodates an integrated and efficient topology of a sensor since bulky horn antennas are not employed. Furthermore, the integration of radio frequency components onto a planar circuit board arrangement having the antenna also disposed thereon dramatically improves manufacturability.  
         [0033]    Sensor  500  further includes at least one controller/signal processing circuit board  530  having a first side  532  for disposing signal processing component  534  thereon and a second side having an electrically conductive ground layer  538 . Electrically conducted ground layer  538  functions as an RF shield when it is oriented in parallel and facing multi-layer radio frequency circuit board  520  upon final assembly within housing  502 . Ground layer  538  also functions as a ground plane for the controller/signal processing circuit board. Signal processing board  530  and radio frequency circuit board  520  interact via connectors  540  and  542 , respectively.  
         [0034]    Sensor  500  further comprises an absorber  550  located between multi-layer radio frequency circuit board  520  and signal processing board  530 . Absorber  550  comes into proximity of both the electrically conductive ground layer  538  on board  530  and the first side  524  having RF components  522  thereon of radio frequency circuit board  520 . In order to minimize the disturbance of the desired electromagnetic fields in the RF structures about and interconnecting RF components  522 , channels or cutouts preferably extending only partially into absorber  550  are incorporated within absorber  550  that provide clearance around such RF components including transmission lines located on first side  524  of radio frequency circuit board  520 .  
         [0035]    [0035]FIG. 6 illustrates an exemplary layout of second side or antenna side  528  of multi-layer radio frequency circuit board  520 . The antenna system of the present invention is placed on the same circuit board as the other RF circuitry. The antenna system includes two coplanar loop series-fed array antennas  570  and  572  that are preferably located on opposite ends of RF circuit board  520  and on the opposite side of the circuit board from RF circuitry components  522 . Thus, in the preferred embodiment, one side of the circuit board includes antennas  570  and  572  with possibly DC interconnect lines  574  miscellaneous bias networks  576  while side  524  includes RF components  522 .  
         [0036]    [0036]FIG. 7 illustrates a detailed layout of a planar printed circuit board antenna, in accordance with a preferred embodiment of the present invention. The present invention utilizes a coplanar waveguide loop antenna  580  for radiating and receiving microwave signals projected about a vehicular target. The present configuration employs a series of radiating elements. In the present implementation, areas of metal are surrounded by slots or areas with no metal. This structure can be implemented by surrounding the metal with air, or by printing it on a dielectric substrate, or on a conductor backed dielelectric substrate. This is a non-resonant element that exhibits many desirable properties such as wide bandwidth, and a low coupling between adjacent elements. The radiation from this element is polarized horizontally as oriented in FIG. 7. In a conductor backed implementation, the elements radiate only out of the page.  
         [0037]    [0037]FIG. 7 depicts an array of series fed coplanar loop elements  582 - 587  and  588 - 593 . In the present invention, the coplanar waveguide traveling wave series loop antenna element exhibits the broadband qualities of a traditional coplanar loop and can be combined in a series like the series fed microstrip patch. For an appreciation of related structures, the following articles are incorporated herein by reference: H. C. Liu, T. S. Horng, and N. G. Alexopoulos, “Radiation of Printed Antennas with Coplanar Waveguide Feed,” IEEE Trans. Antennas Propgat., vol. 43 no. 10, pp. 1143-1148, October 1995; and A. G. Derneryd, “Linearly Polarized Microstrip Antennas,” IEEE Trans. Antennas Propogat., pp. 846-851, November 1976.  
         [0038]    In the present invention, since the coplanar series loop is not a resonant element, the size of the element can be readily adjusted. This size adjustment results in an alteration to the amount of radiation exhibited. Thus, tapered arrays can be designed by utilizing radiating elements with varying sizes. The coplanar waveguide series loop element is implemented using the following features: a conductor backed dielectric substrate  704 ,  702  (see FIG. 9), grounding vias  594  to prevent substrate propagation, and wide coplanar slots  595  to reduce the effects of manufacturing variations.  
         [0039]    The use of a conductor backed dielectric substrate limits the radiation from the element to only one side of the element and also facilitates manufacturing as the element can be printed on a dielectric laminated with metal on both sides. The grounding vias  594  prevent the propagation of parallel plate modes that may exist when dielectric is laminated on both sides by metal. These parallel plate modes could cause coupling between radiating elements printed on the same substrate and could cause unpredictable antenna input impedances.  
         [0040]    The wide coplanar slots  595  help in several ways. First, wide coplanar slots increase radiation and increase tolerance to manufacturing variations. Second, circuit boards are often coated with solder mask and conformal coating to protect the board and components. These coatings, however, fill the coplanar slots and cause unpredictable phase shifts. The widening of the coplanar slots reduces this effect. By way of example and not limitation, slots  595  in the preferred embodiment assume a width of 60 mils. which provides the needed tolerance to manufacturing and coating variations but also maintains the necessary coplanar properties. The width of the center conductor of the coplanar waveguide is chosen to achieve the desired transmission line characteristic impedance.  
         [0041]    The detail of FIG. 7 illustrates vias  594 , that in a preferred embodiment are plated with copper to create a grounded short between the top ground plane and the bottom ground plane. In the present example, vias  594  are placed so that the edge of the via is 25 mil from the edge of the slot which is sufficient to result in marginal influence to the transmission line characteristics but sufficiently close to effectively channel the electromagnetic energy.  
         [0042]    A tapered antenna array  596 ,  597  may be implemented through the use of varying element sizes. Series-fed arrays, such as this one in the present example, are used to replace corporate feed designs in which each element is fed by its own individual transmission line. The corporate feed approach requires an intricate feed structure that becomes more complicated when different antenna elements are used in the array or when a tapered feed is desired. Furthermore, corporate feed structures are prone to undesired radiation which results in antenna pattern distortion.  
         [0043]    The exemplary loop dimensions given on FIG. 7 illustrate the heights of the loops and tapering towards the edges of the array. The radiating edges of the loops are the vertical sides (as oriented in FIG. 7). Thus, variations in the height of the loop results in changes to the degree of radiation from the loop. Consequently, the tapering of the loop size results in a radiation power distribution that creates a radiation pattern with low side lobes. The dimensions shown on the array  597  illustrate the loop widths increasing towards the edges of the array. This ensures that the radiating edges of the loop are in-phase. This dimension results in the widths being larger as the heights are smaller. The exact loops dimensions were determined through simulation.  
         [0044]    The lengths of transmission lines between the loops illustrated on array  596  are adjusted to facilitate every loop radiating in phase. As shown, these lengths are longer for smaller loops. These lengths are again determined from simulation. As illustrated in FIG. 7, the left half of the array  596  is fed from the right and the right half of the array  597  is fed from the left. This would cause an 180° phase shift between the two sides of the array  580  if not compensated for. Since the transmission lines which feed the two sides of the array are of different lengths, the line as illustrated on the right is exactly ½ wavelength longer than the line on the left. This provides the compensation to achieve in-phase radiation from both sides of the array.  
         [0045]    In the present example, the antenna  580  is fed from a  50 Ω transmission line that drives two  100 Ω lines, which intersect at a tee. From the tee to the edges of the array, the transmission lines are  100 Ω. Notice that the  50 Ω transmission line feeding the antenna narrows for a section  598  and then returns to the standard width. This section  598  of the line is a quarter-wave matching section used to provide an impedance match to antenna  580 . The ends of the array are terminated by short-circuited transmission lines. This termination causes a standing wave pattern throughout the antenna and causes the antenna as a whole to become a resonant structure. This has an advantage over a matched termination in that the antenna gain is higher since there are no losses in the termination. If a higher bandwidth antenna is needed, however, a matched termination, which would result in a traveling wave antenna, may be employed.  
         [0046]    [0046]FIG. 8 is a functional block diagram of the radio frequency circuit board with other blocks of related functionality, in accordance with a preferred embodiment of the present invention. The functionality of radio frequency circuit board  520  (FIG. 5) may be partitioned into a transmit portion  602 , including a digitally generated modulated signal generator  603 , and a received portion  604 . Transmit portion  602  is comprised, in the preferred embodiment, of a direct digital synthesizer (DDS)  606  for creating a signal  612  that sweeps in frequency.  
         [0047]    While the present embodiment depicts frequency generation using a DDS, it is also contemplated that other waveform generating devices, generally herein known as digitally generated modulated signal generators, including numerically controlled devices, may be employed for generating effective waveforms In the preferred embodiment, a modulated signal is generated digitally and is thus phase-locked to a digital clock. This modulated signal is then up-converted, if necessary, to the desired band.  
         [0048]    Various embodiments for the digital generation are depicted in FIGS.  8 , and  10 - 12 . FIG. 10 illustrates another embodiment for digitally generating a modulated signal. In this approach, a digitally generated modulated signal generator  780  is comprised of a modulated signal digital generator  782  and an optional up-convertor  784 . In this embodiment, the modulated signal generator provides significant advances over analog signal generators by providing enhanced phase stability over time and improved modulation control which results in lower compression sidelobes and improved detection algorithms.  
         [0049]    [0049]FIG. 11 illustrates another digitally generated modulated signal generator  800 . This embodiment illustrates up-converting a digitally generated modulated signal by using a frequency mixer  802 . In this approach, the modulated signal  804  is mixed with an RF tone  806  resulting in a signal containing frequencies of the sum and difference of the tone and the original signal. Only the sum or the difference frequencies are desired and one or the other must be filtered out by a filter  808 . For example, the digitally modulated signal generator produces a signal ranging from 100 MHz to 150 MHz. This signal is then mixed with a 10.4 GHz tone. The resulting signal contains copies of the digitally generated modulated signal in the 10.3 GHz to 10.25 GHz range and in the 10.5 to 10.55 GHz ranges. A band pass filter  808  with a high Q can be used to filter the lower frequency copy and the higher frequency copy is then transmitted.  
         [0050]    [0050]FIG. 12 illustrates a specific embodiment for digitally generating a modulated signal wherein the digitally generated modulated signal generator  820  is comprised of a direct digital to analog conversion generator  822  capable of direct generation of the desired signal. FIG. 12 further illustrates another implementation of the transmit and receiver portions wherein they share a single antenna that is multiplexed using a circulator  824  for alternating between transmit and receive modes of operation.  
         [0051]    Each of these embodiments comprises similar additional components and the preferred embodiment as illustrated in FIG. 8 is used to describe and define those components. In the preferred embodiment as illustrated in FIG. 8, reference signal  612  sweeps in frequency from 10.5 megahertz to 10.55 megahertz and is generally linear with a duration of 1.25 milliseconds followed by recovery time.  
         [0052]    The output of DDS  606  couples to a phase lock loop  608  which operates by comparing two input frequencies  612 ,  614  and generates a voltage  616  which controls a voltage controlled oscillator (VCO)  610 . Regarding phase lock loop  608 , if the reference signal  612  is lower in frequency than the pre-scaler output  614 , then the output voltage  616  of phase lock loop  608  becomes lowered. Conversely, if reference signal  612  is higher than pre-scaler output  614 , then output voltage  616  of phase lock loop  608  is increased.  
         [0053]    VCO  610  outputs a signal  618  whose frequency is determined by the input voltage  616 . Those of skill in the art appreciate that the higher the input voltage of input  616 , the higher the frequency of the RF signal output  618 , and conversely, the lower input voltage  616 , the lower the frequency of the RF output signal  618 . In a “reverse” drive VCO a change in input voltage yield the opposite result just described. By way of example and not limitation, the VCO  610  of the present embodiment generates an output signal in the 5.25 GHz to 5.275 GHz range.  
         [0054]    Transmit portion  602  is further comprised of a pre-scaler  620  which operates as a frequency divider by reducing the frequency of VCO  610  by a factor of, for example,  4 . Before comparing the two signals, the PLL further divides the signal by a factor of  250  which results in a signal in the 10.5 MHz to 10.55 MHz range, which range is near the same frequency as reference signal  612  as output by DDS  606 . Thus, output signals  612 , from the direct digital synthesizer and pre-scaler output  614  become tracking signals for comparison by phase lock loop  608 . In general, phase lock loop  608  adjusts input voltage  616  to VCO  610  until both inputs, reference signal  612  and pre-scaler output  614 , are at the same frequency. As referenced signal  612  from DDS  606  increases in frequency, phase lock loop  608  drives VCO  610  in such a manner as to also increase the frequency. Thus, output signal  618  from VCO  610  results in the same signal as reference signal  612  other than signal  618  is scaled, in the present example, by a factor of  500 .  
         [0055]    Transmitter portion  602  further includes a Wilkinson divider  622  for dividing the RF signal  618  into two paths while maintaining isolation between the two outputs, output  624  and output  626 . Those of skill in the art appreciate that Wilkinson divider  622  is a splitter in which each output path is reduced by half or  3 dB from input signal  618 .  
         [0056]    Transmitter portion  602  further includes a doubler  628  for receiving signal  624  and generating a signal  630 . Doubler  628  operates as a nonlinear device for effectively doubling the frequency from input signal  624  to output signal  630 . In the present example, input signal  624  operates between 5.25 GHz and 5.275 GHz generating an output  630  ranging from 10.5 GHz to 10.55 GHz. Therefore, signal  630 , in the present example, results in a multiplication of reference signal  612  by a factor of 1,000.  
         [0057]    Transmitter portion  602  further includes an amplifier  632  for coupling with signal  630  and for generating signal  634 . Amplifier  632  provides gain control of the signal for boosting the signal to a level sufficiently large for transmission. Amplifier  632  further couples to a Wilkinson divider  636  for partitioning a portion of the transmission power to the receiver portion through a signal  638  and Wilkinson divider  636  further generates an output  640  for passing to band pass filter  642 . Those of skill in the art appreciate that that pass band filter  642  filters the output signal on the transmit portion to reduce transmissions outside of the desired frequency band. Transmit portion  602  further includes a transmit antenna  644  further described below for emanating the signals generated by the aforementioned circuitry.  
         [0058]    Received portion  604  is comprised of various components for receiving reflected signals as emanated by transmit portion  602 . Reflected signals are received by receive antenna  650  and processed by a bandpass filter  652  which reduces transmission outside of the desired frequency band. The receive filtered signal  654  is thereafter passed to amplifier  656  which generally is implemented as a low noise amplifier for boosting the received signal to a more useable level for processing.  
         [0059]    Amplified signal  658  and signal  638  are received by mixer  660  which, in the present example, is implemented as a nonlinear device that effectively multiplies the two input signals to produce output signal  662 . Those of skill in the art appreciate that mixers operate, for example, by receiving two sinusoidal signals which may be of different frequencies which results in an output signal having the characteristics of the sum of the two input sinusoidal signals, which trigonometrically results in a first frequency corresponding to the sum of the two input frequencies and a second frequency corresponding to the difference of the two input frequencies. This principle is illustrated by the trigonometric identity: sin α cos β=½[sin(α−β)+sin(α+β)] 
         [0060]    Thus, if one input signal is 10.5GHz and a second is 10.50001 GHz then the output signal from the mixer will be the sum of the sinusoids at 21.00001 GHz and another at 10 KHz for the present exemplary implementation, the resulting difference frequency signal is employed for evaluation of the signal characteristics.  
         [0061]    It should be appreciated that the utilization of the difference frequency is a result of ranging capabilities of a linearly sweeping transmitted frequency. For example, the present embodiment utilizes a signal transmitted that is linearly frequency modulated (e.g. chirp). If the transmitted signal is reflected by a single point source target and is received by the radar and mixed with the same linearly modulated signal, the received signal, which has been delayed in time by the propagation duration to and from the target results in a frequency difference between the two inputs to the mixer since the transmitted signal exhibits a constantly increasing frequency during the phase of the period under evaluation. Therefore, the longer the propagation time to and from the target in question, the larger the frequency difference between the presently transmitted and the received signal. For example, in the present illustration, the linearly increasing frequency increases at a rate of 50 MHz in 1.25 milliseconds. Such a linear change in frequency results in a 40 GHz per second change in frequency. Therefore, if a target is located at a distance of 100 feet, the propagation time to and from the target is approximately 203 nanoseconds. In that length of time, the transmit frequency would have changed by 8.13 KHz.  
         [0062]    Received portion  604  is further comprised of a low pass filter  664  which eliminates undesired RF signals from the mixer output, therefore resulting in audio frequencies being present at signal  666 . Therefore, signal  666 , which is the output of the low pass filter  664 , is an audio frequency signal whose frequency corresponds to the range of the target and whose amplitude corresponds to the reflectiveness of the target.  
         [0063]    Receiver portion  604  further includes audio filtering and amplification as illustrated in block  668 . Such filtering and amplification conditions the signal prior to digitization to reduce any feed-through from the transmitting antenna directly coupling to the receiving antenna. Signal conditioning in the form of high pass filtering is employed since transmitter coupling appears in the received signal as a low frequency.  
         [0064]    The following digital circuitry components may reside on a separate digital board. The output condition signal  670  is input to analog-to-digital conversion for  672 , which converts the audio frequency signal to a digital signal for processing and analysis. The digitized output signal  674  is thereafter processed by detection algorithm  676 , which performs spectral analysis on the digitized signal  674  and generates the desired traffic statistics for use in traffic analysis, control, and forecasting. Other processing within detection algorithm  676  include automatic and continuous background estimation, automatic and continuous lane allocation and automatic and continuous detection threshold determination.  
         [0065]    [0065]FIG. 9 illustrates a typical layout of the RF component side of the RF circuit board, in accordance with the preferred embodiment of the present invention. As discussed, RF components  522  (FIG. 5) are populated on side  524 . The transmit portion  602  and receive portion  604  are depicted, absent antennas  570  and  572  which populate the other side of the board. The conductor backed dielectric substrates  704  and  702  for the antenna structures are depicted in FIG. 9. Also depicted in FIG. 9 are the signal via to the transmit antenna  704  and the signal via from the receive antenna  706 .  
         [0066]    The present invention may be embodied in other specific forms without departing from its spirit or essential characteristics. The described embodiments are to be considered in all respects only as illustrative and not restrictive. The scope of the invention is, therefore, indicated by the appended claims rather than by the foregoing description. All changes which come within the meaning and range of equivalency of the claims are to be embraced within their scope.