Abstract:
The invention relates to a method and to an electronic circuit for clipping of at least first CS 1  and second input CS 2  signals to provide an output signal which does not exceed a predefined threshold, the electronic circuit comprising input means for inputting of a first sample of the first input signal and of a second sample of the second input signal, means  12  for applying a criterion on the first and second samples, means for clipping the first and/or second sample if the criterion is fulfilled in order to enable subsequent filter and summation operations of the first and second input signals such that the predefined threshold is not exceeded.

Description:
FIELD OF THE INVENTION  
         [0001]    The present invention relates to the field of clipping and, more particularly, to a method and an electronic circuit for mitigating signal distortions in a communication system.  
         BACKGROUND AND PRIOR ART  
         [0002]    It is as such known from the prior art to use clipping techniques to prevent amplifier saturation. Clipping is a useful technique especially for signals with a high peak to average power ratio (PAPR). Such signals are used in systems which use multi-code CDMA or multi sub-channel OFDM. Such systems need output power amplifiers with large dynamic ranges.  
           [0003]    If the amplifiers cannot handle the peak powers then the resulting saturation causes intermodulation products and adjacent channel interference (ACI). A number of techniques exist for reducing the peak to average ratio, but many of these are modulation dependent, for example, coding, partial transmit sequences, phasing for OFDM and Multi-Carrier Spread Spectrum.  
           [0004]    WO 98/44668 shows a method for reducing the PAPR of a composite carrier signal. A peak-reducing waveform is estimated and summed with a composite signal to reduce a peak-to-average power ratio of the composite signal. The estimate of the peak-reducing waveform is modified to have Walsh code components orthogonal to the assigned Walsh codes. An iterative process of estimating subsequent peak-reducing waveform is implemented to produce a peak-reducing waveform which, when summed with the composite signal, results in a composite signal having a peak-to-average ratio at a desired level and thus does not have the effects of remodulating the assigned Walsh codes. Constraints on the magnitude of the unassigned Walsh code components can be included for controlling the power level under the unassigned Walsh codes.  
           [0005]    EP 545 596 A1 shows a deviation limiting transmission circuit which comprises a soft clipper which performs measurements at two nodes. The soft clipper limits both its own output signal and the output signal of a low-pass/band-stop filter to selected maximum values, thus preventing prolonged deviation overshoots.  
           [0006]    EP 106 7683 A1 shows a clipping method whereby the clipping is performed dependent on succeeding filtering taking into account the filter characteristics. This method is only applicable for single carrier systems.  
           [0007]    Further an adaptive technique is known which reduces the risk of overcompensating (overclipping). This is achieved by using an additional filter to predict the response of the pulse-shaping filter from which the exact amount of compensation can be calculated.  
           [0008]    It is an object of the present invention to provide an improved clipping method and electronic circuit for clipping, in particular for direct sequence code division multiple access (DS-CDMA) cellular telecommunication systems.  
         SUMMARY OF THE INVENTION  
         [0009]    The underlying problem of the present invention is solved basically by applying the features laid down in the independent claims. Preferred embodiments of the invention are given in the dependent claims.  
           [0010]    The present invention is particularly advantageous for multi-carrier amplifiers. The invention enables to perform a joined clipping operation rather than clipping each carrier separately. Clipping is performed when a criterion is fulfilled which is common to all signals to be amplified by the multi-carrier amplifier.  
           [0011]    This is particularly advantageous as a joined clipping criterion can prevent unnecessary clipping. For example clipping can be avoided in accordance with the present invention in a situation where one of the signals has a strong amplitude while signals intended for other carriers have a low amplitude as the overall signal will not exceed the threshold level for the linear range of the amplifier.  
           [0012]    In accordance with a preferred embodiment of the invention the criterion is the sum of the amplitudes of the current signal samples being input into the clipper. If the sum of the amplitudes is below a predefined threshold level no clipping is performed. If the sum of the amplitudes exceeds the predefined threshold level clipping is performed for example by multiplying all of the sample amplitudes with a factor which is smaller than one.  
           [0013]    In accordance with a further preferred embodiment of the invention the subsequent pulse shaping filtering and frequency multiplexing of the signals is taken into consideration for the clipping operation. This way re-growth of the clipped peaks above the threshold level caused by interpolation in the pulse shaping filter is avoided. Another advantage is that the phase angles of the samples can also be considered for the clipping in addition to the amplitude. This has the advantage that unnecessary clipping is avoided, especially in situations where one or more of the samples are of opposite or substantially different phase angles. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0014]    In the following preferred embodiments of the invention are described in greater detail by making reference to the drawings in which:  
         [0015]    [0015]FIG. 1 is a block diagram of a first embodiment of a system for predictive clipping for a multi carrier solution,  
         [0016]    [0016]FIG. 2 is an alternative embodiment where the interpolation and frequency multiplexing is taken into consideration for the clipping,  
         [0017]    [0017]FIG. 3 is illustrative of the generation of the frequency multiplexing signals,  
         [0018]    [0018]FIG. 4 is a block diagram of a preferred embodiment of the clipper of FIG. 2,  
         [0019]    [0019]FIG. 5 is illustrative of the transformation formulas used within the clipper of FIG. 4. 
     
    
     DETAILED DESCRIPTION  
       [0020]    [0020]FIG. 1 shows a part of an electrical circuit  10  of a transmitter of a radio base station of a DS-CDMA cellular telecommunication system. The electrical circuit  10  has an number of summers  11 , which are coupled to a clipper  12 .  
         [0021]    The clipper  12  is connected to a number of pulse shaping filters  13  which are in turn connected to multipliers  14  for frequency conversion. The outputs of the multipliers  14  are added by means of summer  15 . The output of summer  15  is coupled to digital-to-analogue-converter  16  which serves to convert the digital signal outputted by summer  15  to an analogue signal which is to be amplified by means of an amplifier which is not shown in FIG. 1.  
         [0022]    In operation each of the summers  11  is coupled to a number of channels C 1 , C 2 , . . . Cn. The information of each of the channels C 1 , C 2 , . . . Cn belonging to the same summer  11  is summed in that summer  11  to generate a respective composite signal CS 1 , CS 2 , . . . CSn.  
         [0023]    Each of the composite signals CS 1 , CS 2 , . . . CSn is inputted to the clipper  12  which produces the clipped composite signals A 1 , A 2 , . . . , An. Each of the clipped composite signals A 1 , A 2 , . . . , An is inputted into the corresponding pulse shaping filter  13 . This results in the signals B 1 , B 2 , . . . , Bn, respectively. The clipped and filtered composite signals B 1 , B 2 , . . . , Bn are inputted into respective multipliers  14  for frequency conversion. The frequency converted signals B 1 , B 2 , . . . , Bn are summed by means of summer  15  to create a frequency multiplexed multi carrier signal C.  
         [0024]    The clipper  12  determines the amplitude of all of the composite signals CS 1 , CS 2 , . . . , CSn. Further the clipper  12  generates an internal signal by summing up all of these amplitudes. The total of the amplitude values is than compared to a predefined threshold value. If the total of the amplitudes is below the threshold value no clipping is performed. This means that the output signals A 1 , A 2 , . . . , An are equal to the input signals CS 1 , CS 2 , . . . , CSn. In other words the clipper  12  is transparent when no clipping is performed.  
         [0025]    When the total of the amplitude values exceeds the predefined threshold value clipping is performed by the clipper  12 . In this instance a factor is calculated by the clipper  12  by dividing the threshold value by the total of the amplitude values. This factor is by definition smaller than one. Then the clipper  12  multiplies all of the composite signals CS 1 , CS 2 , . . . , CSn with the factor in order to clip the corresponding signal samples. This results in clipped output signals A 1 , A 2 , . . . , An.  
         [0026]    The joined clipping reduces effectively the peak to average ratio obtained on the multi carrier signal C which is provided to the amplifier. This results in a better power efficiency of the amplifier.  
         [0027]    [0027]FIG. 2 shows an alternative embodiment, where like elements are designated by the same reference numerals as in the embodiment of FIG. 1.  
         [0028]    The electrical circuit  17  of FIG. 2 has an LO-generator  18  for generating a signal p 1  for the frequency conversion L 1  as well as a LO-generator  19  for generating a signal p 2  for frequency conversion L 2  within the multipliers  14 , respectively. The signals p 1  and p 2  are inputted into delay elements  20  and  21 , respectively for delaying the signals p 1  and p 2 . By means of the delayed signals p 1  and p 2  the frequency conversions L 1  and L 2  are carried out in the multipliers  14 , respectively.  
         [0029]    Further the signals p 1  and p 2  are inputted into the demultiplexer  22  and  23 , respectively.  
         [0030]    The demultiplexer  22  outputs the sub-signals p 11 , p 12 , p 13  and p 14  of the signal p 1  and the demultiplexer  23  outputs the sub-signals p 21 ,p 22 , p 23 , p  24  of the signal p 2 . These demultiplexed signals are inputted into the clipper  12 .  
         [0031]    In the example considered here the clipper  12  receives at its input the input signals S 1  and S 2  corresponding to the composite signals CS 1  and CS 2  of FIG. 1. The input signals S 1  and S 2  are processed within clipper  12  by means of the demultiplexed sub-signals and by means of the filter coefficients of the pulse shaping filters  13 .  
         [0032]    As the number of input signals S 1  and S 2  is equal to n=2 in the example considered here an over-sampling coefficient of at least four is required. The pulse shaping filter  13  has a length of M. For the purposes of the clipper  12  the pulse shaping filter  13  is approximated by a filter of length 7 with the filter coefficients a, b, c, d, c, b, a. These coefficients are at the same time the central coefficients of the pulse shaping filters  13 . In the example considered here the pulse shaping filters  13  are identical; however it is important to note, that this is not essential and that the pulse shaping filters for the different channels can have different filter lengths and/or filter coefficients.  
         [0033]    The operation of the clipper  12  is predictive as it involves the subsequent interpolations performed by the pulse shaping filters  13  and the frequency conversions L 1  and L 2 . This is made possible by providing the sub-signals of the signals p 1  and p 2  to the clipper  12  and by providing a priori knowledge to the clipper  12  regarding the characteristics of the pulse shaping filters  13 . The delay elements  20  and  22  are necessary in the preferred embodiment of FIG. 2 to account for the delay caused by the processing within the clipper  12  and the delay caused by the pulse shaping filters.  
         [0034]    The two LO-generators  18  and  19  generate complex signals with amplitudes equal to one and with a phase dependent on the frequency conversion L 1  or L 2 . The output signals p 1  and p 2  are sampled at four times chip speed. The demultiplexing of the signals p 1  and p 2  into the four separate signals, respectively, is performed in a “Round Robin” way, as illustrated in FIG. 3 with respect to the signal P 1 .  
         [0035]    [0035]FIG. 4 shows a block diagram of an embodiment of the clipper  12 . The clipper  12  has a module  24  for calculating a value H 4  by means of a function f 4  having parameters S 11 , S 21 , d, p 14  and p 24 .  
         [0036]    The filter coefficient d of the pulse shaping filters  13  is present in the module  24  as a priori knowledge. The signal S 11  is equal to the input signal S 1  and the signal S 21  is equal to the input signal S 2 . Both input signals S 11  and S 21  are inputted into the module  24  as well as the sub-signals p 14  and p 24  (cf. signals p 1  and p 2  of FIG. 2 and FIG. 3).  
         [0037]    Further the module  24  has the value of the threshold T as a priori knowledge. The absolute value of H 4  is compared to the threshold value T. If the absolute value of H 4  exceeds the threshold value T then a factor Y 1  is calculated. The factor Y 1  is calculated by dividing the threshold T by the absolute value of H 4 . If the absolute value of H 4  does not exceed the threshold value T the factor Y 1  is equal to one by definition.  
         [0038]    The factor Y 1  is outputted from the module  24  and inputted into the multipliers  25  for multiplication of the input signals S 1  and S 2  with Y 1 . This results in the signals S 12  and S 22 , respectively.  
         [0039]    The signals S 12  and S 22  as well as the sub-signals p 11 , p 12 , p 13  and p 21 , p 22 , p 23  are inputted into the module  26 . The module  26  serves to calculate values H 1 , H 2  and H 3 .  
         [0040]    The value of H 1  is a function f 1  of the signals S 12 , S 22 , the filter coefficients a and c, the sub-signals p 11  and p 21  as well as the further signals S 13  and S 23 . The value of H 2  is determined by means of the function f 2  which has the parameters S 12 , S 22 , S 13 , S 23 , b, p 12  and p 22 . The value of H 3  is determined by means of the function f 3  having the parameters S 12 , S 22 , c, S 13 , S 23 , a, p 13  and p 23 .  
         [0041]    The module  26  determines the maximum of the absolute values of H 1 , H 2  and H 3  which is the value H. If H exceeds the threshold value T then the scaling factor Y 2  equals T divided by H. If the contrary is the case the scaling factor Y 2  is equal to 1.  
         [0042]    The factor Y 2  is outputted by the module  26  and inputted into the multipliers  27 ,  28  and  29 ,  30 , respectively. The other input of the multiplier  27  is the signal S 12  which is multiplied by Y 2 .  
         [0043]    The output of the multiplier  27  is inputted into the delay element  31 . The output of the delay element  31  is the input of the multiplier  28  which provides the output signal A 1 . The output of the delay element  31  is at the same time the signal S 13  which is inputted into the module  26 .  
         [0044]    The input of the multiplier  29  is the signal S 22  which is multiplied by the factor Y 2 . The output of the multiplier  29  is inputted into the delay element  32 . This provides the output S 23  which is inputted into the module  26  and into the multiplier  30  for multiplication with the factor Y 2 . The output of the multiplier  30  is the output signal A 2 .  
         [0045]    [0045]FIG. 5 shows the functions f 1 , f 2 , f 3  and f 4  for calculating H 1 , H 2 , H 3  and H 4 , respectively.