Abstract:
A method for compensating a transceiver for impairments includes transmitting a plurality of partial bandwidth training signals using a transmitter. A plurality of response signals of a receiver having a bandwidth and exhibiting receiver impairments is captured. Each response signal is associated with one of the partial bandwidth training signals. Each of the partial bandwidth training signals is associated with a portion of the receiver bandwidth. A plurality of partial compensation filters is generated based on the plurality of response signals. Each partial compensation filter is associated with one of the response signals. The partial compensation filters are combined to configure a receiver compensation filter operable to compensate for the receiver impairments.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     Not applicable. 
     BACKGROUND 
     The disclosed subject matter relates generally to telecommunication and, more particularly, to a method and apparatus for compensating for transceiver impairments. 
     In telecommunication, direct-conversion techniques involve mixing a baseband signal with a carrier signal without using an intermediate frequency. A direct conversion receiver (DCR), also known as homodyne, synchrodyne, or zero-IF receiver, may then demodulate incoming signals by mixing the incoming signal with a local oscillator signal synchronized in frequency to the carrier signal. The baseband signal may then be obtained simply by low-pass filtering the mixer output, without requiring further detection. 
     A direct conversion transceiver may be implemented on a single chip, making it inexpensive and versatile. However, the transmission and receive paths for the base band signals (i.e., in-phase (I) and quadrature (Q)) are independently formed. Small variances in these paths introduce amplitude and phase variations in the respective transmit and receive signals, otherwise referred to as IQ imbalance impairments. 
     To calibrate a direct conversion transceiver, the impairments must be identified and compensated for prior to use. This compensation is difficult because it is necessary to calibrate the direct conversion transmitter and direct conversion receiver individually using a separate external reference for each path. If an uncompensated receiver is used to calibrate a transmitter over the complete frequency span, the transmitter calibration will be corrupted by the impairments of the receiver, and vice versa. 
     This section of this document is intended to introduce various aspects of art that may be related to various aspects of the disclosed subject matter described and/or claimed below. This section provides background information to facilitate a better understanding of the various aspects of the disclosed subject matter. It should be understood that the statements in this section of this document are to be read in this light, and not as admissions of prior art. The disclosed subject matter is directed to overcoming, or at least reducing the effects of, one or more of the problems set forth above. 
     BRIEF SUMMARY 
     The following presents a simplified summary of the disclosed subject matter in order to provide a basic understanding of some aspects of the disclosed subject matter. This summary is not an exhaustive overview of the disclosed subject matter. It is not intended to identify key or critical elements of the disclosed subject matter or to delineate the scope of the disclosed subject matter. Its sole purpose is to present some concepts in a simplified form as a prelude to the more detailed description that is discussed later. 
     One aspect of the disclosed subject matter is seen in a method for compensating a transceiver for impairments. The method includes transmitting a plurality of partial bandwidth training signals using a transmitter. A plurality of response signals of a receiver having a bandwidth and exhibiting receiver impairments is captured. Each response signal is associated with one of the partial bandwidth training signals. Each of the partial bandwidth training signals is associated with a portion of the receiver bandwidth. A plurality of partial compensation filters is generated based on the plurality of response signals. Each partial compensation filter is associated with one of the response signals. The partial compensation filters are combined to configure a receiver compensation filter operable to compensate for the receiver impairments. 
     Another aspect of the disclosed subject matter is seen a transceiver including a transmitter, a receiver, a receiver capture unit, a receiver compensation filter, and a receiver compensation estimation unit. The transmitter is operable to transmit a plurality of partial bandwidth training signals. The receiver has impairments and is operable to receive the plurality of partial bandwidth training signals. The receiver capture unit is operable to capture a plurality of response signals of the receiver. Each response signal is associated with one of the partial bandwidth training signals. Each of the partial bandwidth training signals is associated with a portion of the receiver bandwidth. The receiver compensation filter is operable to filter signals received by the receiver. The receiver compensation estimation unit is operable to generate a plurality of partial compensation filters based on the plurality of response signals. Each partial compensation filter is associated with one of the response signals. The receiver compensation estimation unit is operable to combine the partial compensation filters to configure the receiver compensation filter to compensate for the receiver impairments. 
    
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
       The disclosed subject matter will hereafter be described with reference to the accompanying drawings, wherein like reference numerals denote like elements, and: 
         FIG. 1  is a simplified block diagram of a transceiver in accordance with one illustrative embodiment of the present subject matter; 
         FIGS. 2A-2C  illustrate training signals used for compensating a receiver in the transceiver of  FIG. 1 ; 
         FIG. 3  is a diagram modeling receiver impairments for the transceiver of  FIG. 1 ; and 
         FIG. 4  is a diagram of a full bandwidth training signal for compensating a transmitter in the transceiver of  FIG. 1 . 
     
    
    
     While the disclosed subject matter is susceptible to various modifications and alternative forms, specific embodiments thereof have been shown by way of example in the drawings and are herein described in detail. It should be understood, however, that the description herein of specific embodiments is not intended to limit the disclosed subject matter to the particular forms disclosed, but on the contrary, the intention is to cover all modifications, equivalents, and alternatives failing within the spirit and scope of the disclosed subject matter as defined by the appended claims. 
     DETAILED DESCRIPTION 
     One or more specific embodiments of the disclosed subject matter will be described below. It is specifically intended that the disclosed subject matter not be limited to the embodiments and illustrations contained herein, but include modified forms of those embodiments including portions of the embodiments and combinations of elements of different embodiments as come within the scope of the following claims. It should be appreciated that in the development of any such actual implementation, as in any engineering or design project, numerous implementation-specific decisions must be made to achieve the developers&#39; specific goals, such as compliance with system-related and business related constraints, which may vary from one implementation to another. Moreover, it should be appreciated that such a development effort might be complex and time consuming, but would nevertheless be a routine undertaking of design, fabrication, and manufacture for those of ordinary skill having the benefit of this disclosure. Nothing in this application is considered critical or essential to the disclosed subject matter unless explicitly indicated as being “critical” or “essential.” 
     The disclosed subject matter will now be described with reference to the attached figures. Various structures, systems and devices are schematically depicted in the drawings for purposes of explanation only and so as to not obscure the disclosed subject matter with details that are well known to those skilled in the art. Nevertheless, the attached drawings are included to describe and explain illustrative examples of the disclosed subject matter. The words and phrases used herein should be understood and interpreted to have a meaning consistent with the understanding of those words and phrases by those skilled in the relevant art. No special definition of a term or phrase, i.e., a definition that is different from the ordinary and customary meaning as understood by those skilled in the art, is intended to be implied by consistent usage of the term or phrase herein. To the extent that a term or phrase is intended to have a special meaning, i.e., a meaning other than that understood by skilled artisans, such a special definition will be expressly set forth in the specification in a definitional manner that directly and unequivocally provides the special definition for the term or phrase. 
     Referring now to the drawings wherein like reference numbers correspond to similar components throughout the several views and, specifically, referring to  FIG. 1 , the disclosed subject matter shall be described in the context of a transceiver  100  including a transmit path  110 , a receive path  120 , and a sampling receive path  130 . The transmit path  110  and the sampling receive path  130  are connected to a directional coupler  140 , which is in turn connected to a filter panel  150  and an antenna  160 . The directional coupler  140  prevents reflections from the filter panel  150  from affecting the sampling receive path  130 . The filter panel  150  (i.e., commonly referred to as a duplexer) separates the outgoing transmit signal from the incoming receive signal, performs bandpass filtering of the transmit RF output, and performs bandpass filtering of the receiver RF input. 
     The transmit path  110  includes a transmitter  111  including a modulator  112  and a digital-to-analog converter (DAC)  113 , a transmitter IQ compensation filter  114 , a transmitter IQ compensation estimation unit  115 , a transmitter capture unit  116 , a power amplifier  117 , and a switch  118  for selecting between normal data traffic or a training signal  119 . The receive path  120  includes a receiver  121  including a demodulator  122  and an analog-to-digital converter (ADC)  123 , a receiver IQ compensation filter  124 , a receiver IQ compensation estimation unit  125 , and a receiver capture unit  126 . The sampling receive path  130  includes a sampling receiver  131  including a demodulator  132  and an ADC  133 , a sampling receiver IQ compensation filter  134 , a sampling receiver IQ compensation estimation unit  135 , and a sampling receiver capture unit  136 . The transmitter  111 , receiver  121 , and sampling receiver  131  are implemented using hardware, while the various units illustrated in  FIG. 1  are implemented in the digital portion of the transceiver  100  (i.e., by a processing unit executing software or firmware). Hence, the units generally describe functions rather than discrete hardware. However, depending on the particular implementation, dedicated hardware may be employed to perform one or more of the functions attributed to the units. 
     The transmitter  111 , receiver  121 , and sampling receiver  131  employ inphase (I) and quadrature (Q) signals. The separate paths for these components are not shown for ease of illustration. Generally, the I and Q paths are separate, such that amplitude and phase variations exist between the paths. These variations cause separate impairments in the transmitter  111 , the receiver  121 , and the sampling receiver  131 . The transmitter  111  and the sampling receiver  131  operate in the same frequency band. In the illustrated embodiment, the sampling receiver  131  is employed to compensate for transmitter impairments. After the transmitter  111  is compensated, it may be used to generate a training signal for compensating the receiver  121 . However, because the sampling receiver  131  is used to estimate and compensate the transmitter  111 , the impairments of the sampling receiver  131  must first be estimated and compensated. As known to those of ordinary skill in the art, the sampling receiver  131  may also be employed for functions such as digital predistortion or time division duplexing (i.e., functioning as the only receiver). 
     For a typical IQ transmitter  111 , the unimpaired baseband signal may be represented by:
 
 s ( t )= s   I ( t )+ js   Q ( t ),  (1)
 
where s I (t), s Q (t) are the baseband in-phase and quadrature signal components.
 
     The output of the modulator  112  is given by:
 
 x ( t )= i ( t )+ jq ( t ),  (2)
 
where
 
 i ( t )=α I   [s   I ( t )cos(φ I )+ s   Q ( t )sin(φ I )]
 
 q ( t )=α Q   [s   Q ( t )cos(φ Q )+ s   I ( t )sin(φ Q )]′,  (3)
 
and α I ,α Q ,φ I ,φ Q  are the modulator imbalance errors (i.e., impairments).
 
     Equation (2) can be rewritten in the form of a summation of two components comprised of a desired signal and its conjugate,
 
 x ( t )=α 1   s ( t )+α 2   s *( t ),  (4)
 
where
 
     
       
         
           
             
               
                 
                   
                     
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     The complex conjugate of x(t) is given by:
 
 x *( t )=α 1   *s *( t )+α 2   *s ( t ),  (6)
 
and by using x(t) and x*(t) as separate observations, the compensation required to minimize the imbalance errors in α I ,α Q ,φ I ,φ Q  can be determined by exploiting the conjugate symmetry nature of the IQ imbalance. For purposes of illustration, the imbalance errors are provided here as scalar errors, but generally, the imbalance errors are functions of frequency α I (f),α Q (f),φ I (f),φ Q (f).
 
     To perform the compensation analysis, this symmetrical nature is employed by generating partial bandwidth training signals and measuring the full bandwidth response signal. Signal components arising from impairments are detected in the portion of the bandwidth not covered by the partial bandwidth training signal. The partial bandwidth process is continued until the full bandwidth has been covered, and the response signals from the partial bandwidth iterations are combined to compensate the receiver over the full bandwidth. For purposes of the following examples, a half bandwidth technique is used (i.e., two iterations), however, it is contemplated that more than two partial bandwidth partitions may be employed. 
     To facilitate compensation, the transmitter  111  is configured to select the training signal  119  using the switch  118  rather than the normal data traffic.  FIG. 2A  illustrates an exemplary transmit signal  200  that covers half of the calibration bandwidth. In the illustrated embodiment, the signal  200  is an arbitrary wideband signal that need not be flat with respect to frequency.  FIG. 2   b  shows the response of the receiver  131  to the transmit signal  200 . The actual transmit signal  210  transmitted by the modulator  112  and received by the receiver  131  is shown in  FIG. 2B . Note that a transmitter impairment signal  220  is also present due to the IQ imbalances of the transmitter  111 . The transmit signal  210  shown in  FIG. 2B  is transmitted by setting the local oscillator frequency of the modulator  112  to have a negative center frequency offset by approximately one fourth of the calibration bandwidth (i.e., −BW/4) from the center frequency of the sampling receiver  131 , which places the transmit signal  210  into the lower half bandwidth of the sampling receiver  131 . Due to the symmetrical nature of the IQ imbalance, the resulting receiver impairment signal  230  occupies the upper half-bandwidth and is separated from the transmitter impairment signal  220 . Note that the receiver impairment signal  230  includes an impairment signal  240  representing impairments of the sampling receiver  131  arising from the transmit signal  200  and a receiver impairment signal  250  representing the impairments of the sampling receiver  131  arising from the transmitter impairment signal  220 . 
     A second iteration is performed to cover the second half of the calibration bandwidth, as shown in  FIG. 2C . The transmit signal  210 ′ (i.e., including a transmit signal  200 ′ and a transmitter impairment signal  220 ′) is transmitted by setting the local oscillator frequency of the modulator  112  to have a positive center frequency offset by approximately one fourth of the calibration bandwidth (i.e., +BW/4) from the center frequency of the sampling receiver  131 , which places the transmit signal  210 ′ into the upper half bandwidth of the sampling receiver  131 . Again, due to the symmetrical nature of the IQ imbalance, the resulting receiver impairment signal  230 ′ occupies the lower half-bandwidth and is separated from the transmitter impairment signal  220 ′. The receiver impairment signal  230 ′ includes an impairment signal  240 ′ representing impairments of the sampling receiver  131  arising from the transmit signal  200 ′ and a receiver impairment signal  250 ′ representing the impairments of the sampling receiver  131  arising from the transmitter impairment signal  220 ′. 
     For each partial bandwidth iteration, the sampling receiver capture unit  136  records the response signal. Each partial bandwidth capture is used to generate a compensation filter for the corresponding bandwidth. The sampling receiver IQ compensation estimation unit  135  uses the capture information to generate coefficients for the sampling receiver IQ compensation filter  134 . 
       FIG. 3  illustrates an exemplary model  300  of the sampling receiver  131  impairments. The impairments are modeled by coefficients h 1 , h 2  in blocks  310 ,  320 . The modeled impairments h 1 , h 2  are added into the desired signals s 1 [n], s 2 [n] by adders  330 ,  340 , resulting in the impaired baseband signals x 1 [n], x 2 [n]. The compensation conducted by the sampling receiver IQ compensation estimation unit  135  generates filter coefficients w 1 , w 2  in blocks  350 ,  360 , which are subtracted from the impaired baseband signals in adders  370 ,  380  to attempt to remove the impairments, resulting in received signals ŝ 1 [n], ŝ 2 [n]. The decorrelation criteria block  390  represents the convergence criteria for the decorrelation algorithm. When the decorrelation criteria are met, the iterative adjustment of the filter taps is terminated. 
     As indicated above, the baseband signal x(t) can be written as the summation of the desired signal s(t) and its conjugate s*(t). The conjugate component after the down conversion process appears as an image opposite DC in the baseband frequency domain. To perform a compensation, the conjugate contribution in x(t) may be minimized though a blind adaptive estimation of a 1  and a 2  shown in Equation (5). One exemplary method to separate mixed signals is to use second order statistics to observe and minimize the correlation between the sources, provided that the source s 1 (t)=s(t) and its conjugate s 2 (t)=s*(t) are uncorrelated. The signals are assumed to be zero mean, complex Gaussian random processes. The correlation for these signals is given by:
 
 C   ss *(τ)= E[s ( t +τ)( s *( t ))*].  (6)
 
     Separating in phase and quadrature components in the correlation of Equation (6), results in:
 
 C   ss *(τ)=[ C   s     I     s     I   (τ)− C   s     Q     s     Q   (τ)]+ j[C   s     I     s     Q   (τ)+ C   s     Q     s     I   (τ)]  (7)
 
     The imaginary part of the correlation is zero if the in-phase and quadrature components of the signal are uncorrelated. The real part of the correlation is zero if the autocorrelation of both in-phase and quadrature components are the same. Although in-phase and quadrature component autocorrelation is not exactly the same for typical 3G and 4G broadband signals, the correlation is small, such that minimization through an adaptive scheme is possible. 
     The decorrelation of the received signals ŝ 1 [n], ŝ 2 [n] requires that C ŝŝ *[0]=E[ŝ 1 [n]ŝ* 2 [n]]=0. This condition further leads to 
                         C       s   ^     ⁢       s   ^     *         ⁡     [   0   ]       =         +     (       h   2   *     -     w   2   *       )       ⁢     (     1   -       w   1     ⁢     h   2         )     ⁢       R       y   1     ⁢     y   1         ⁡     [   0   ]         +       (       h   1     -     w   1       )     ⁢     (     1   -       w   2   *     ⁢     h   1   *         )     ⁢       R       y   2     ⁢     y   2         ⁡     [   0   ]             ,           (   8   )               
where the mixing coefficients are given by
 
                 h   1     =         b     a   *       ⁢           ⁢   and   ⁢           ⁢     h   2       =       b   *     a         ,         
and R y     1     y     1   , R y     2     y     2    are autocorrelation coefficients of y 1  and y 2 , respectively. The mixing coefficients are conjugates of each other, which implies w 2 =w 1 *. Assuming a simple case with only one tap to estimate, this is readily accomplished with a straightforward minimization algorithm that is similar to a Newton zero search, where the zero is in the cross correlation domain.
 
     In a simple example, the sampling receiver IQ compensation estimation unit  135  may use the following signal separation procedure. 
     1. Capture signal x[n] into sampling receiver capture unit  136 . 
     2. Form x 1 [n]=x[n] and x 2 [n]=x*[n]. 
     3. Form Signal Estimates.
 
 ŝ   1   [n]=x   1   [n]−w   1   *[n]x   2   [n] 
 
 ŝ   2   [n]=x   2   [n]−w   2   *[n]x   1   [n].   (9)
 
     4. Update the coefficient weights. 
                         w   1     ⁡     [     n   +   1     ]       =         w   1     ⁡     [   n   ]       +     2   ⁢           ⁢   μ   ⁢         s   ^     2     ⁡     [   n   ]       ⁢         s   ^     1   *     ⁡     [   n   ]             ⁢     
     ⁢         w   2     ⁡     [     n   +   1     ]       =       w   1   *     ⁡     [     n   +   1     ]         ⁢     
     ⁢       0   &lt;   μ   &lt;     1       R       x   2     ⁢     x   2         ⁡     [   0   ]           ,             (   10   )               
where μ is a convergence parameter for the LMS algorithm, and R x     2     x     2    is an autocorrelation coefficient of x 2 .
 
     The IQ imbalance observed on typical commercial modulators and demodulators is frequency dependent, and a single tap solution is not usually adequate to compensate the IQ imbalance image across a wide bandwidth. To accommodate wide bandwidth compensation, the expressions in equations (6) through (10) can be vectorized generate a compensation filter of length 2L+1,
 
 W   (n)   =[w   (n) (− L ) . . .  w   (n) ( L )] T   ,−L≦k≦L,   (1)
 
where
 
 w   1   (n+1) ( k )= w   1   (n) ( k )+2μ( ŝ   2   [n]ŝ   1   *[n−k ]),− L≦k≦L  
 
 w   2   (n+1) ( k )= w   1   (n) ( k )*.
 
     If a solution exists to decorrelate ŝ 1 [n] and ŝ 2 *[n], then
 
 C   ŝŝ   *[k]=E[ŝ   1   [n]ŝ*   2   [n−k]]= 0,− L≦k≦L   (2)
 
over the filter span 2L+1. This implies that the filter response h 1 =w 1 ,h 2 =w 2 . Although similar in appearance to an LMS form of adaptive equalization, this approach uses the cross correlation in a minimization search.
 
     The steps described above consider the filters W 1  and W 2  to be odd symmetric, or conjugate filters. Real world demodulators are not symmetric, and exhibit varying frequency response on either side of the receiver center frequency. 
     To adequately compensate a demodulator using the blind methods outlined above, the sampling receiver IQ compensation estimation unit  135  uses a multiple step approach to build partial bandwidth filters (e.g., two half-band filters). The partial bandwidth filters are then spliced together in the frequency domain to form the final full band compensation filter. 
     To generate the full bandwidth filter coefficients for the sampling receiver IQ compensation filter  134 , the sampling receiver IQ compensation estimation unit  135  converts the time-domain coefficients of each partial bandwidth filter to the frequency domain by a discrete Fourier transform (DFT). The partial bandwidth response signals are joined in the frequency domain to form a frequency response that is defined over the entire receiver bandwidth. At intersection points (e.g., f=0 for two half-bandwidth filters) the responses are averaged to define the midpoint of the full-bandwidth response. Then an inverse DFT is performed to convert the full-bandwidth response back to time domain coefficients. 
     After the sampling receiver IQ compensation filter  134  has been configured using the partial BW captures to build a full BW compensation filter for the sampling receiver  131 , the transmitter  111  can be calibrated by using the compensated sampling receiver  131  as a training reference. As shown in  FIG. 4 , the training signal  119  is now configured as a full bandwidth signal  400  including a transmit signal  410  and a transmitter impairment signal  420 . However, compensation of the transmitter  111  is possible now that the sampling receiver  131  may be used as a compensated training reference. 
     The subsequent final compensation of the transmitter  111  may be conducted using conventional techniques known to those of ordinary skill in the art. For example, the full bandwidth training signal is captured by the transmitter capture unit  116  prior to being modulated by the modulator  112  (i.e., prior to the transmitter impairments being introduced by the modulator  112 ). The signal received by the sampling receiver  131  is compensated by the sampling receiver IQ compensation filter  134  to remove the sampling receiver impairments. The filtered signal is then provided to the transmitter IQ compensation estimation unit  115 , which compares the compensated received signal to the transmit signal captured by the transmitter capture unit  116 . Prior to transmitter compensation, the compensated received signal includes transmitter impairments. By comparing the compensated received signal to the transmit signal, the transmitter IQ compensation estimation unit  115  identifies the impairments and generates coefficients for the transmitter IQ compensation filter  114  to compensate for the transmitter impairments. For clarity and to avoid obscuring the present subject matter, these conventional techniques are not described in greater detail herein. 
     The techniques described herein allow compensation of the transmitter  111  using the compensated receiver  131  without the use of an external training signal. This approach simplifies the compensation techniques, thereby potentially reducing the complexity and cost of the transceiver. 
     The particular embodiments disclosed above are illustrative only, as the disclosed subject matter may be modified and practiced in different but equivalent manners apparent to those skilled in the art having the benefit of the teachings herein. Furthermore, no limitations are intended to the details of construction or design herein shown, other than as described in the claims below. It is therefore evident that the particular embodiments disclosed above may be altered or modified and all such variations are considered within the scope and spirit of the disclosed subject matter. Accordingly, the protection sought herein is as set forth in the claims below.