Abstract:
A traffic sensor is mounted at a fixed location to monitor multiple lanes of traffic. The traffic sensor (a) generates a programmable time-varying modulating signal; (b) generates a modulated microwave signal based on the programmable time-varying modulating signal; (c) radiates the modulated microwave signal in a radiation beam at an object; (d) provides a proportional calibration signal based on the modulated microwave signal; (e) measures parameters of the calibration signal, and (f) corrects the programmable time-varying modulating signal based on the parameters of the calibration signal.

Description:
FIELD OF THE INVENTION 
   The present invention relates in general to traffic sensors, and more particularly relates to a traffic sensor and method for providing a stabilized signal. 
   BACKGROUND OF THE INVENTION 
   As urban centers increase in size, and traffic congestion becomes more common, the need for accurate and up-to-date traffic information also increases. Traffic surveillance relies primarily on traffic sensors, such as inductive loop traffic sensors that are installed under the pavement. Alternatively, video sensors may also be used to obtain traffic information. 
   Residing underground, inductive loop sensors are expensive to install, replace and repair because of the associated roadwork required. Moreover, such roadwork also causes traffic disruptions. Video sensors, on the other hand, are cheaper, but have other drawbacks, such as an inability to operate in the dark or in weather that impairs visibility, such as fog or snow. 
   To overcome these drawbacks, radar sensors have been employed to obtain traffic information. Radar sensors typically transmit low-power microwave signals at the traffic, and detect vehicles based on the reflected signals. Radar sensors are generally cheaper than inductive loop traffic sensors, and, unlike video sensors, operate in the dark and in a wide range of weather conditions. 
   For proper and stable operation, frequency modulated continuous wave (FMCW) radars must transmit a stable and preferably linear frequency sweep. Any non-linearities of the sweep can reduce range resolution. Further, changes in the df/dt sweep slope due to temperature drift may reduce the accuracy of ranges measured, and shifts in the center frequency due to temperature changes can push the transmission signal out of the FCC allocated band. 
   One solution to this problem involves digitally synthesizing the radar transmit signal such that at all times it is derived to be some numerical multiple of a fixed low crystal-controlled reference frequency; however, this approach tends to be expensive both in terms of production cost and power consumption. 
   Another approach adopted is to rely on an analog oscillator to generate the transmit signal. In this case the frequency modulation would be performed by a single varactor diode. The main drawbacks of this approach are non-linearity and temperature drifts. One solution is to provide a frequency generation oscillator (FGO) circuit with a memory chip. This memory would be loaded with numbers defining a function derived from the polynomial N(x)=Ax 2 +Bx+C in which the A coefficient denotes the non-linear part, the B coefficient denotes the primary modulation slope and the C coefficient relates to the center frequency. 
   By testing each transceiver after manufacture, its A, B and C coefficients can be determined and recorded in the memory resulting in a linearized sweep, and substantially correct slope and center frequency as required. Further, through a combination of real-time temperature measurement and statistical analysis of batches of microwave transceivers, temperature correction can be applied to the A, B and C coefficients so as to stabilize the sweep function and center frequency by compensating for variation in these coefficients due to temperature. However, despite this compensation, there may be substantial drift of the df/dt slope. Accordingly, this approach while being cheaper both in terms of cost and power consumption than the above-described digital solution, may suffer from the drawback of reduced accuracy. 
   SUMMARY OF THE INVENTION 
   In accordance with an aspect of the present invention there is provided a vehicular traffic sensor for mounting at a fixed location to monitor multiple lanes of traffic. The vehicular traffic sensor comprises: (a) a processor unit for generating a programmable time-varying modulating signal; (b) a transceiver unit for receiving the programmable time-varying modulating signal from the processor unit and for generating a modulated microwave signal based on the programmable time-varying modulating signal; (c) at least one antenna for (i) receiving the modulated microwave signal from the transceiver unit, (ii) forming a radiation beam, (iii) radiating the modulated microwave signal in the radiation beam at an object; and, (iv) receiving the modulated microwave signal reflected back from the object; and (d) a signal stabilizing unit for receiving a portion of the modulated microwave signal from the transceiver unit and for providing a proportional calibration signal back to the processor. The processor unit is further operable to (i) receive the calibration signal from the signal stabilizing unit, (ii) measure parameters of the calibration signal, and (iii) correct the programmable time-varying modulating signal based on the parameters of the calibration signal. 
   In accordance with a second aspect of the present invention, there is provided a method of operating a traffic sensor for mounting at a fixed location to monitor multiple lanes of traffic. The method comprises: (a) generating a programmable time-varying modulating signal; (b) generating a modulated microwave signal based on the programmable time-varying modulating signal; (c) radiating the modulated microwave signal in a radiation beam at an object; (d) providing a proportional calibration signal based on the modulated microwave signal; (e) measuring parameters of the calibration signal, and (f) correcting the programmable time-varying modulating signal based on the parameters of the calibration signal. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     A detailed description of preferred embodiments is provided herein below with reference to the following drawings, in which; 
       FIG. 1 , in a block diagram illustrates a traffic sensor in accordance with an aspect of the present invention; 
       FIG. 2 , in a block diagram illustrates a transceiver unit and a signal stabilizing unit of the traffic sensor of  FIG. 1 ; 
       FIG. 3 , in a block diagram illustrates a processor unit of the traffic sensor of  FIG. 1 ; and, 
       FIG. 4 , in a flowchart, illustrates a method of determining a non-linear curve representing a signal to be fed to the transceiver unit to provide a substantially linear output signal from the transceiver unit in accordance with a further aspect of the invention. 
   

   DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
   Referring to  FIG. 1 , there is illustrated in a block diagram a traffic sensor  20  in accordance with an aspect of the present invention. The traffic sensor  20  comprises a processor unit  22 , a transceiver unit  24 , an antenna  26  and a signal-stabilizing unit  28 . The processor unit  22  generates a programmable time-varying modulating signal  30  that is received by the transceiver unit  24 . The transceiver unit  24  takes the programmable time-varying modulating signal and generates a modulated microwave signal  32  that is received by the antenna  26 . The antenna  26  then forms a radiation beam and radiates the modulated microwave signal  32  in the radiation beam  34  at an object or objects, such as a vehicle (not shown) or several vehicles. The antenna  26  also receives back from the object a reflected modulated microwave signal  36 . 
   The signal-stabilizing unit  28  receives a calibration portion  38  of the modulated microwave signal  32 . The signal-stabilizing unit  28  then derives a proportional calibration signal  40 , which is transmitted back to the processor unit  22 . The processor unit  22  is further operable to measure parameters of the calibration signal  40  and to correct the programmable time varying modulating signal  30  based on the parameters of the calibration signal  40 . A more detailed description of the operation of the transceiver unit  24 , signal stabilizing unit  28  and processor unit  22  is provided below. 
   Referring to  FIG. 2 , the transceiver unit  24  and the signal-stabilizing unit  28  are illustrated in more detail in a block diagram. As shown, the transceiver unit  24  comprises a receiver unit  24 A and a transmitter unit  24 B. The transmitter unit  24 B receives the programmable time-varying modulator signal  30  and transmits the modulated microwave signal  32 . The receiver unit  24 A receives the reflected modulated microwave signal  36  from the antenna  26 . 
   The transmitter unit  24 B comprises a voltage tunable dielectric resonator oscillator (VT-DRO)  42 , which outputs a microwave signal  46  whose frequency is determined by the programmable time-varying modulating signal  30  input to the VT-DRO  42 . The transmitter unit  24 B further comprises a power divider  48  that divides the microwave signal  46  into two signals  46   a  and  46   b.  Typically the power divider  48  is a circuit that accepts an input signal and delivers multiple output signals equal in phase and in a fixed amplitude ratio. 
   The generation of microwave signal  46  by the VT-DRO  42  may introduce unwanted distortions at specific frequencies. To remove these unwanted distortions from the modulated microwave signal  32  provided to the antenna  26 , a notch filter  50  is included in the transmitter unit  24 B. The notch filter  50  receives the divided signal  46   a  and produces modulated microwave signal  32  by passing through most of the frequencies of divided signal  46   a  unaltered, while attenuating those in a narrow range to very low levels. In one embodiment, the notch filter  50  is a 21 GHz distributed notch filter selected to remove harmonic distortions introduced to the transmitted signal  46   a  by the VT-DRO  42 . 
   The receiver unit  24 A receives the reflected modulated microwave signal  36  from the antenna  26 . The receiver unit  24 A comprises a mixer  52 , which effectively multiplies divided signal  46   b  from power divider  48  and reflected modulated microwave signal  36  together to generate a mixed signal  54 . The mixed signal  54  is at the base band, in that it is determined as the difference in frequency between the reflected modulated microwave signal  36  and the signal  46   b,  which difference in frequency represents the time elapsed between transmission and reception of the modulated microwave signal. This time elapsed, in turn, represents the distance to the object from which the signal was reflected. The mixed signal  54  is received by an amplifier  56 , which produces an amplified signal  58 . The amplified signal  58  is then sent to the processor unit  22  as illustrated in  FIG. 1 . 
   The signal-stabilizing unit  28 , which is also shown in more detail in  FIG. 2 , comprises various components for down-converting the modulated microwave signal  46  received from the VT-DRO  42  to a lower baseband signal suitable for digital processing by the processor unit  22 . In a specific embodiment illustrated in  FIG. 2 , a phase-locked voltage controlled oscillator (PLL VCO)  60  is locked to a clock signal  62  with a frequency f 1  and outputs an adjusted clock signal having frequency f 2 =nf 1 /m, where n and m are integers typically called frequency divider parameters. In one embodiment, the PLL VCO  60  is an Analog Devices ADF4360 and is configured to produce an output signal of 2624.72 MHz from an input clock signal of 24.4912 MHz. 
   The adjusted clock signal  64  output by the PLL VCO  60  is provided to a frequency multiplier  66 , which receives the adjusted clock signal  64  with frequency f 2  and produces a reference signal  68  having a frequency of Nf 2 . In other words, the frequency multiplier  66  takes the adjusted clock signal  64  and produces an output signal at the N th  harmonic of the frequency of the input signal  64 . In one embodiment the frequency multiplier  66  is a Hittite Microwave HMC443LP4 Quadrupler and produces a 10.4988 GHz reference signal from a 2624.72 MHz input signal. 
   The signal-stabilizing unit  28  further comprises a mixer  70  for mixing the modulated microwave signal  46  and the reference signal  68  from the frequency multiplier  66 . The mixer  70  multiplies these two signals to produce two output band signals. One of these signals is the up-conversion frequency band, having a frequency equal to the sum of the frequencies of the two input signals, while the other output signal is at the down-conversion band and has a frequency equal to the difference between the frequencies of the signals  46  and  68 . These two output signals  72  from the mixer  70  are provided to a low pass filter  74 , which passes through the down-conversion signal of the output signals  72 , while filtering out the up-conversion signal of the output signals  72 . In one embodiment, the low pass filter cut-off frequency is 50 MHz such that the low pass filter  74  will pass frequencies lower than 50 MHz, while stopping frequencies higher than 50 MHz. 
   The signal-stabilizing unit  28  further comprises an amplifier  80 , which receives the down-conversion component  78  of the output signal  72  received from the low-pass filter  74 . The amplifier  80  amplifies this down-converted signal  78  to a level sufficient for direct counting by digital circuits, to obtain the proportional calibration signal  40 , which is sent to the processor unit  22  as illustrated in  FIG. 1 . 
   Referring to  FIG. 3 , the processor unit  22  is illustrated in more detail in a block diagram. The elements of the processor unit  22  illustrated in  FIG. 3  correct the programmable time-varying modulating signal  30  to increase the linearity of the modulated microwave signal  32  generated by the transceiver unit  24  based on the programmable time-varying modulating signal  30 . In general, frequency modulated continuous wave radar works most effectively when the modulated microwave signal from the transceiver is linearly modulated. 
   The programmable time-varying modulating signal  30  is generated by a frequency generation oscillator (FGO) circuit  82 . Since the VT-DRO  42  (as shown in  FIG. 2 ) is a non-linear device, to produce a linear output the input to the VT-DRO  42  must be non-linear. The non-linear function required to produce a linear output can be approximated by the function N(x)=Ax 2 +Bx+C, where the A coefficient denotes the non-linear part, the B coefficient denotes the primary modulation slope and the C coefficient relates to the centre frequency. The FGO circuit  82  comprises a memory unit (MU) which stores a fixed number of values of the function N(x)=Ax 2 +Bx+C. 
   The processor unit  22  further comprises a counter  86 , which is used to count from 1 to N and then down from N to 1 where N is the number of function values  88  stored in the memory unit  84 . When the processor unit  22  is running in normal mode, the number generated by the counter  86  is sent to the FGO circuit  82  where it is used to retrieve the function value  88  written at that address of the memory unit  84 . Once retrieved, the function value  88  is sent to a digital-to-analog converter (DAC)  90 , which produces the time-varying modulating signal  30  based on the function value  88  received from the memory unit  84 . 
   While the processor unit  22  generally runs in normal mode, at a fixed interval it switches into calibration mode. According to one embodiment the calibration mode interval is set at 3 minutes. In calibration mode, the system runs the curve-fitting process illustrated in the flowchart of  FIG. 4 . The curve-fitting process involves determining three points on the curve represented by the function N(x)=Ax 2 +Bx+C, and then using those three values to determine the coefficients, A, B and C. To determine a point on the N(x) curve, one has to determine the value of N(x1) and the value of x1. To determine both N(x1) and x1 the Micro Processing Unit (MPU)  92  varies an input signal  94  provided to the DAC  90 . More specifically, a curve fitter function  96  provided by the MPU  92  varies the input signal  94  provided to the DAC  90  until the frequency of the output signal  46  from the VT-DRO  42  (see  FIG. 2 ) reaches a pre-determined frequency. At that point, the value of the input signal  94  will be N(x1) and the x1 value is calculated from the desired frequency. Subsequently, the MPU  92  will know where on the ideal linear curve that the desired frequency sits. Thus, the MPU  92  will know that it has to use the N(x1) value at the equivalent time in its cycle, which is x1. 
   Referring to  FIG. 4 , there is illustrated in a flow chart a curve-fitting process in accordance with a further aspect of the invention. The curve-fitting process begins with step  98  in which the curve fitter  96  of the MPU  92  sends signal  94  to the DAC  90  in an attempt to generate frequency f 1 . Based on the signal  94 , the DAC  90  generates the programmable time-varying modulating signal  30 , which is received by the transceiver unit  24  as shown in  FIG. 1 . As described above in connection with  FIG. 2 , the proportional calibration signal  40  is derived from the output signal  46  from the VT-DRO  42 . Referring back to  FIG. 3 , this proportional calibration signal  40  is fed back to an analog-to-digital converter (ADC)  100 , which generates an equivalent digital signal  102 , which is transmitted to a frequency counter  104 , which reads the frequency of the signal  102 . This frequency reading is equal to the sum of the frequency of the signal  102  from the ADC  100  and the frequency of the reference signal  68  ( FIG. 2 ). This calculated transmit frequency is then compared to f 1 . If the calculated transmit frequency is not equal to f 1  (within a selected tolerance), then query  108  of the method of  FIG. 4  returns the value “NO” and the curve fitter  96  of the MPU  92  returns to step  98 . When, however, the calculated transmit signal is equal to f 1  then query  108  returns the value “YES” and the curve fitter  96  of MPU  92  proceeds to step  110  where the DAC input value  94  that produced the f 1  is stored as NO. 
   In step  112 , the curve fitter  96  adjusts value  94  sent to the DAC  90  in an attempt to generate a second frequency f 1 +Δf, which is now the desired frequency. If the calculated transmit frequency, determined as described above, is not equal to the desired frequency, then the query  114  returns the value “NO” and the curve fitter  96  returns to step  112  to adjust DAC input value  94 . When, however, the calculated transmit signal is equal to the desired frequency, query  114  returns the value “YES” and the process proceeds to step  116  in which the DAC input value  94  that produced the desired transmit frequency is stored as N 1 . 
   In step  118  the curve fitter  96  once again adjusts DAC input signal  94  in an attempt to generate a frequency of f 1 +2Δf, which is now the desired frequency. If the calculated transmit frequency, determined as described above, is not equal to the desired frequency, then the query  120  returns the value “NO” and the curve fitter  96  returns to step  118  to adjust the DAC input value  94 . If, on the other hand, the calculated transmit signal equals the desired frequency then query  120  returns the value “YES” and the curve fitter  96  proceeds to step  122  where the DAC input value  94  that produced the desired transmit frequency is stored as N 2 . 
   Once N 0 , N 1  and N 2  have all been determined in steps  110 ,  116  and  122  respectively, the curve fitter  96  solves for coefficient A, B and C from the N 0 , N 1  and N 2  values in step  124 . Then, if the coefficients result in a polynomial N(x) that is different from the one that had been used during the previous normal operation mode, then the new N(x) is used to generate new N(x) values, and the memory unit  84  is updated with the new N(x) values. The manner in which the processor unit  22  switches between the normal mode of operation and the calibration mode of operation is described in more detail below. 
   As described above, the counter  86  of the processor unit  22  is used to count from 1 to N and then down from N to 1 where N is the number of function values stored in the memory unit  84 . In one embodiment, it takes very little time—say half a millisecond—for the memory unit to count up from 1 to N, or down from N to 1. This feature of the processor unit  22  is important in the switch between the normal mode and calibration mode of operation in this embodiment as described below. 
   According to one embodiment, the processor unit  22  comprises a clock  126 , which in addition to providing the clock signal  62  described above in connection with  FIG. 2 , also determines whether a pre-defined calibration mode interval such as, say 3 minutes, has elapsed. If this calibration mode interval has elapsed, then during the half millisecond in which the counter  86  is counting down from N to 1 the curve-fitter  96  will temporarily interrupt the signal  88  sent from the memory unit  84  to the DAC  90 , and instead transmit the signal  94  directly from the curve fitter  96  to the DAC  90 . This signal  94  is adjusted as described above to calibrate the programmable time-varying modulating signal  30  output by the DAC to linearize the modulated microwave signal  32  from the transceiver unit  24 . It will, of course, be apparent to those of skill in the art that calibration might occur more frequently or less frequently or alternatively might proceed concurrently with the normal mode of operation by simply monitoring the particular proportional calibration value  40  provided by a particular signal  88  from the MU during the normal mode of operation. 
   Preferably, the memory unit  84  comprises two distinct memories. After the curve-fitter  96  has determined N 0 , N 1  and N 2  in steps  110 ,  116 , and  120  respectively, and is solving for the polynomial function N(x) in step  124 , one component of the memory unit  84  can continue to supply values  88  to the DAC  90 . In fact, this component of memory unit  84  can continue to supply values  88  to the DAC  90  even while the other component of the memory unit  84  is updated to include the new values determined using the new polynomial functions N(x). Then, when all of the new values have been stored, the memory unit can be immediately switched from one component of the memory to the other component of the memory in which the new values of the polynomial N(x) are stored. The other memory component can then be used to store the values of the next polynomial function N(x) determined during the next calibration mode. 
   Other variations and modifications of the invention are possible. All such modifications or variations are believed to be within the sphere and scope of the invention. As defined by the claims appended hereto.