Abstract:
Circuits for correcting base line shift of the detector coupling circuit of a TOFMS provide gain and impedance characteristics that compensate for the AC coupling effect of the detector. In one circuit, base line correction is achieved by injecting a current equal to that which flows due to the buildup charge in the detectors AC coupling network. In another circuit, the current source drives an integrator which is coupled to the signal path to reduce the detector AC coupling effects. In another circuit, a low noise amplifier utilizes a feedback network that reduces the detector AC coupling effects. In yet another circuit, an operational amplifier is employed to reduce the detector AC coupling effects.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application claims priority under 35 U.S.C. §119(e) on U.S. Provisional Application No. 60/878,299 entitled Base Line Restoration Circuit, filed on Jan. 3, 2007, by Ted J. Casper, the entire disclosure of which is incorporated herein by reference. 
    
    
     BACKGROUND OF THE INVENTION 
     The present invention relates to circuits that compensate for the base line offset caused by AC coupling of a detection system. 
     Ion detectors suitable for time-of-flight mass spectroscopy (TOFMS) have become available that allow the data collection system to be at a different potential than the ion detecting device. This allows the ion detector voltages to be optimized for sensitivity and resolution without requiring the output stage of the ion detector to be near ground potential. These detectors use a DC blocking capacitor to couple the electron current from the detector output to the data collection system. An example of this type of detector would be the ETP AF882123. This particular detector uses a 1000 pF high voltage capacitor to couple the detector output to the connector which feeds the data collection system. A second 1000 pF capacitor provides a return path for the current and is connected between the detectors output bias supply and ground. The value of this capacitance may be different for various detector models and manufacturers. In addition, the capacitance value will vary with applied detector bias voltages. 
     Although this method provides high frequency coupling, the lower frequency components are not transferred to the data collection system resulting in a base line shift this is dependant on output magnitude and duration. This issue has been recently acknowledge by ETP (a division of SGE Analytical Science Pty. Ltd.) in an Application note titled “ETP Electron Multiplies MagneTOF™ Detector Applications Notes Preliminary Version—13/9/06”. 
     Typical ion detectors output current pulses that are Gaussian in shape. These pulses are defined by their peak amplitude and the width of the pulse at one half peak amplitude. This width is commonly referred to as Full Width Half Height (FWHH) in the scientific community. Typical output pulses from an ion detector can range from a FWHH of 400 pS to greater than 15 nS with amplitudes ranging from micro volts to hundreds of mili volts when measure at 50 Ohms. Historically, the last grid or plate in the detector was terminated into a ground referenced 50 Ohm system. The 50 Ohm system allows for easy connection from the detector to the data collection system using conventional coaxial cable. 
     DC coupling of a detector may not be possible due to high voltages present on the detection device. An example is an electron multiplier or micro channel plate used for the detection of ions or other charged particles. The AC coupling circuit creates an average offset in the base line that is equal to the average output current of the detector times the detectors output load resistance. In addition, detector pulses create instantaneous offsets that are dependent on the height and width of the pulses, thereby creating a base line that is peak dependant. 
     SUMMARY OF THE INVENTION 
     This invention compensates for such instantaneous base line shifts. The present invention provides methods and circuits for correcting base line shift by providing gain and impedance characteristics that compensate for the AC coupling effect of the detector. In one embodiment, the circuit achieves base line correction by injecting a current equal to that which flows due to the buildup charge in the detectors AC coupling network. In another embodiment, the current source drives an integrator which is coupled to the signal path to cancel or reduce the detector AC coupling effects. In yet another embodiment, a low noise amplifier stage utilizes a feedback network that cancels or reduces the detector AC coupling effects. In yet a further embodiment of the invention, an operational amplifier is employed which cancels or reduces the detector AC coupling effects. 
     In each of these embodiments, the base line restoration for a time-of-flight mass spectrometer is provided to correct for errors introduced by the AC coupling network of an ion detector. 
     These and other features, objects and advantages of the present invention will become apparent upon reading the following description thereof together with reference to the accompanying drawings. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     A computerized model of a typical AC coupling network (modeled in P-Spice) illustrates the effects of the detector AC coupling circuit. The model is shown in  FIG. 1 . The AC coupling section of the ETP detector is inside block  10 . Actual detector output signals would be negative in polarity. Positive pulses were utilized in this simulation to simplify the discussion. 
     Piecewise Gaussian pulses were created that feed current sources (G 1 -G 4 ) to simulate the current output of the detector. To obtain multiple pulses, current sources were summed. Four pulses were injected in the following sequence; 2 mV 2 nS FWHH (full width at half height), 500 mV 15 nS FWHH, 2 mV 2 nS FWHH, 2 mV 2 nS FWHH. The output voltage across R 6  and R 4  to ground was measured. With the +HV resistance at 1 MEG (simulating a filter) a DC shift occurs at R 4 . The magnitude of the shift is equal to 50/5050 of the voltage across R 6  (the sum of the capacitor C 1  and C 4  voltages). The signal time delays are shown by T 1 -T 6 . 
       FIG. 2  shows the simulation results with R 5 =1 MEG. The instantaneous offset produced is equal to the integrated current from the detector times the reciprocal of the capacitance present. For instance, to produce a 0.5V signal peak requires 10 mA into 50 Ohms (the circuit load resistance). The integrated current of a Gaussian 10 mA 15 nS FWHH pulse is 159.9 pA-Seconds. The voltage across 500 pF of capacitance (C 1  in series with C 4 ) would be 319.8 mV which is close to the value calculated. This voltage is divided between R 6  and the impedance across R 4  (including R 4 ). Since the load resistance is 50 Ohms, the DC offset across R 4  created by the single 500 mV, 15 nS FWHH pulse would be 50/5050 time 319.8 mV or 3.16 mV. This offset will discharge at a time constant of the total R times C or in this case 5050 times 500 pF. For the offset to decay to within 500 uV, (¼ of a 2 mV signal) 4.66 uS of signal free time would have to occur. The average voltage across R 6  will be equal to the Detector average output current times 5050 provided the current consists of pulses that occur at a rate more frequent than the time constant of the circuit. If R 5  is reduced to near 0, (direct connection to +HV supply) C 1  is no longer dominant in the circuit, and the offset for a given previous peak becomes smaller due to the increased time constant of 1 nF time 5050 Ohms versus the 1 nF/2 times 5050 ohms of the filtered +HV. The average R 6  voltage will still be equal to the average Detector current times 5 k. 
       FIG. 3  shows simulation results with R 5 =0.01. Base line shift will be present for the ETP AF882123 Detector. The instantaneous shift will be peak dependant and will therefore vary by composition and concentration. Depending on the spectral components of a sample, the offsets could tend to mask trailing mass events by shifting them positive. As noted earlier, the computer model simulations used inverted currents than what the actual detector will produce. 
     These base line shifts can be corrected through many means, including variations to the methods discussed. Modifications of these methods or use of different circuits and/or components could be made without departing from the spirit or scope of this invention. The following are examples of circuits and their operation that can be used to correct for the base line shift effect. 
     EXAMPLE 1 
     Integrate and Inject Compensation Current 
     In one embodiment, the circuit of  FIG. 4  achieves base line correction by injecting a current equal to that which flows due to the built up charge in the detector&#39;s AC coupling network of C 1 , C 4  and R 6 . When the +HV supply is connected to a stiff supply (which is preferable in most cases), C 1  is no longer a factor in the time constant. To mimic the response of R 6  and C 4 , the base line restore circuit  20  of  FIG. 4  employs a low pass filter R 8  and C 5  coupled in shunt with the termination resistance R 4 , the data systems input impedance. The data system could be the input to an analog to digital (A/D) converter or an amplifier stage. C 3  is present to block DC current thus preventing run away of the compensation over time. The value of C 3  is chosen to be much greater than C 5 . R 1  provides a DC reference and has a resistance which is much greater than that of R 8 . The voltage measured on C 5  is converted to a current by voltage controlled current source G 5  and its output added to the current flow at the node of R 4 , C 4  and R 8 . The preferred values of the circuit elements are shown on the circuit diagram of  FIG. 4 . 
     Since the voltage across C 5  is near identical to the offset created across R 4  due to AC coupling, G 5  is adjusted to inject a current that offsets or cancels the AC coupling effect.  FIG. 5  shows the results of the same stimulus used on  FIG. 3 . 
       FIG. 3  shows a base line drift of greater than 1.5 mV. With the base line restore circuit  20 , however, the base line drift is reduced to near zero as shown in  FIG. 5 . Actual implementation of G 5  could be accomplished by utilizing (for example) a transconductance amplifier like the OPA660, OPA860 or OPA615 from Texas Instruments (also for the current sources G 1 -G 5  in all circuits). Transistors or operational amplifiers could also be utilized to create the required current source. This implementation has the draw back of adding wide band noise to the low level signals coming from the detector since the current source directly feeds the load resistance. Noise will be added up to the bandwidth of the amplifier which for the OPA660, OPA860, or OPA615 can reach over 400 MHz. 
     EXAMPLE 2 
     Current Driven Integrator Compensator 
     A noise performance improvement to the integrate and inject compensation circuit of  FIG. 4  can be made by moving the integration function to after the current source. This allows the noise bandwidth of the circuit to be limited by the integration capacitor. A preferred circuit of this type is illustrated in  FIG. 6 . In this circuit  30 , the compensation circuit is connected to the load termination. Transconductance amplifier U 1  creates a current at pin  8  that is proportional to the voltage present at pin  3  divided by the resistance present in the emitter (E) circuit. U 1 &#39;s bias current is controlled by R 5  which adjusts the internal emitter resistance. This internal resistance added to R 3  represents the resistance seen by the emitter circuit for AC signals. In this case, the total AC emitter resistance is adjusted to be 50 Ohms. The DC emitter resistance is much higher eliminating potential lockup of the circuit due to high DC values. The DC emitter resistance is set by R 2 . The value of C 3  is chosen to not affect performance for the widest pulses and amplitudes that must be base line corrected. In this case, the widest pulse of interest was considered to be the 20 nS FWHH. CINT and RFEEDBACK are chosen to match the time constants created by the AC Coupling section of the detector and RLOAD. For optimum cancellation of the AC coupled detector effect, the time constant of CINT and RFEEDBACK will match that of the AC Coupling section of the detector. In a preferred implementation, either RFEEDBACK or CINT would be an adjustable component to allow the system to be optimized for different detector operating voltages. For instance, a typical installation would have CINT set to about 680 pF and RFEEDBACK set to about 5 k. In this application, the BYPASS capacitor is much larger than C 1  making the value of C 2  at its operating voltage the deterministic value. 
     Since the collector current of U 1  is equal to the input voltage divided by 50 Ohms (described above), the current in CINT matches the current flowing through C 2 . The voltage developed across CINT therefore matches the voltage developed across C 2  for a given input pulse from the detector. This developed voltage on CINT is feed back to RLOAD and the detector through RFEEDBACK. Since R 1  equals RFEEDBACK and the voltages across their associated capacitors are equal, the currents of RFEEDBACK and R 1  are equal resulting in no current flow in RLOAD. Both CINT and the C 2  will decay in voltage at the same rate since the time constants are equal. 
     This circuit will add lower frequency noise to the system below the break point set by RFEEDBACK and CINT. In this implementation, added noise would be present below approximately 50 kHz. A variation of this circuit would be to place a large (greater than 100 times CINT) blocking capacitor in series with RFEEDBACK and short C 3 . This variation allows for lower frequency response but results in higher drift of DC bias points in U 1 . 
     EXAMPLE 3 
     Low Noise Amplifier Base Line Correction 
     The previous circuits can be added to an existing data collection system to remove the effects of base line shift caused by AC detector coupling. However, both circuits increase the total system noise due to the added active components. Some data collection electronics include a low noise amplifier circuit to boost the signal level to a value adequate for further analysis. Typical noise levels for this type of preamplifier are specified in the range of 1.5 nV/Hz^0.5. To achieve these noise levels, discrete transistor designs are typically utilized that employ feedback techniques to create the desired 50 Ohm input impedance. Feedback is used since a termination resistor of 50 Ohms will have a noise level of 0.9 nV/Hz^0.5. 
     A typical low noise amplifier  40  ( FIG. 7 ) will consist of a low noise RF transistor Q 1  which is feed through a large blocking capacitor C 1 . Feedback is created by R 4  in the emitter circuit and R 1  in the collector to base circuit. The values shown result in an input impedance of approximately 50 Ohms from 100 Hz through 400 MHz. 
     This circuit  40  can be modified, as seen in  FIG. 8 , by adding a capacitor CFB in series with R 1  to increase the gain and input impedance at a desired frequency. Adding capacitor CFB results in increased low frequency gain and a boost in low frequency input impedance that compensates for the voltage developed across the detector coupling capacitors. It should be noted that some amplifiers employ a blocking capacitor like CFB. However, the intent is to optimize the value of CFB for operation with an AC coupled detector. For the same detector utilized previously, it was found a CFB value of 0.017 uF was optimum to restore the base line of pulses less than 20 nS FWHH without increasing amplifier noise levels. 
     EXAMPLE 4 
     Typical Low Noise Amplifier with Adjustable Base Line Correction 
     In the preferred implementation, CFB is adjustable through electronic or mechanical means. Adjustments can be made by trimmer capacitors, varactor diodes, switches, or relays appropriate for 400 MHz frequencies. An example of a relay that can be utilized at these frequencies is a TT712-68CSP made by TeraVicta. This relay is a MEMS device capable of 7 GHz performance in a 3.5 mm by 4.6 mm package. Utilizing a pair of TT712-68CSP relays results in 16 possible combinations of compensation values as shown in the circuit  50  of  FIG. 9 . A typical installation would have the values of CFB in crease by factors of two resulting in binary capacitance selection. For example, if CFB is 0.012 uF, and 4% step sizes are desired, CFB 1  would be 500 pF, CFB 2  would be 1000 pF, CFB 3  would be 2000 pF, and CFB 4  would be 4000 pF. The total capacitance range available would be 0.012 uF to 0.0195 uF. 
     EXAMPLE 5 
     Operational Amplifier Circuits 
     Correction for the base line shift could also be done utilizing operational amplifiers configured as integrators or low frequency boost circuits if noise level is not of primary concern. One possible solution would be to utilize a current feedback amplifier like the Intersil HFA1130 as shown by circuit  60  in  FIG. 10 . In this design, C 1  and R 4  have a similar time constant as the detector AC coupling network. R 1  and R 6  in parallel set the input impedance of the circuit to 50 Ohms. 
     It will become apparent to those skilled in the art that various modifications to the preferred embodiment of the invention as shown in the above examples and described herein can be made without departing from the spirit or scope of the invention as defined by the appended claims.