Abstract:
A high-frequency phase shifter  1  for varying the phase between its high-frequency input signal and its output signal by the transmission phase Φ, consisting of a two-port network  2  which is symmetrical in relation to input and output and which with respect to its high-frequency properties consists of three two-terminal networks  5  consisting of low-loss reactances  8 , wherein at least one of the two-terminal networks  5  is arranged in a series circuit as a two-terminal network  6  in series with one of the connecting ports  23  and at least one of the two-terminal networks  5  is arranged in a parallel circuit as a two-terminal network  7  in parallel with the two-port earth  9 , so that a symmetrical T-circuit  24  or a symmetrical π circuit  25  is provided.

Description:
TECHNICAL FIELD 
       [0001]    The invention relates to a high-frequency phase shifter  1  for varying the phase between its high-frequency input signal and its output signal by the transmission phase Φ, consisting of a two-port network  2  which is symmetrical in relation to input and output and which with respect to its high-frequency properties consists of three two-terminal networks  5  consisting of low-loss reactances  8 , wherein at least one of the two-terminal networks  5  is arranged in a series circuit as a two-terminal network  6  in series with one of the connecting ports  23  and at least one of the two-terminal networks  5  is arranged in a parallel circuit as a two-terminal network  7  in parallel with the two-port earth  9 , so that a symmetrical T-circuit  24  or a symmetrical π circuit  25  is provided. 
       BACKGROUND OF THE INVENTION 
       [0002]    High-frequency phase shifters are used for example in antenna technology for phase-controlled antennae to swivel the radiation pattern or to shape it. In this case there is usually the requirement to enable swivelling of the radiation pattern electronically. This leads to the demand for a high-frequency phase shifter which is electronically controllable in its transmission phase Φ. 
         [0003]    A phase shifter for electronic variation of the phase is known from DE 3802662A1. This consists of a series phase shifter assembly, the range of variation of the individual phase shifters being small. The phase is in this case varied by means of switched diodes. For continuous tracking of the radiation pattern of an antenna, however, it is desirable to vary the phase continuously, i.e. in analogue fashion, by means of an electrical signal. A phase shifter which is variable in the transmission phase Φ electronically and in analogue fashion is known from DE69127128. The phase shifter consists of a four-terminal hybrid circuit with a branch line connected to only two variable-capacitance diodes or varactors of which the capacitance value is varied by means of a control voltage. A structure of this kind is admittedly able to sweep a large angular range of the phase, but with only two variable-capacitance diodes it does not allow appropriate compensation of the influences, to the effect that, on the input and output sides, adequate reflection loss is achieved in relation to a reference characteristic impedance Z0 on the input and output sides. 
       SUMMARY OF THE INVENTION 
       [0004]    It is therefore the object of the present invention to provide a high-frequency phase shifter of which the transmission phase Φ is electronically controllable and, with ease of implementation, variable in analogue fashion over a wide angular range and with particularly low mismatching. 
         [0005]    The invention is described in claim  1 . The subsidiary claims contain advantageous embodiments and developments of the invention. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0006]    Practical examples of the invention are shown in the drawings and described in more detail below. In detail they show: 
           [0007]      FIG. 1 : basic forms of a phase shifter according to the invention
       a) symmetrical phase shifter with T-structure, formed from the two two-terminal networks  6  in series, consisting of the series circuit comprising the fixed inductance  10  in series and the series capacitance element  12  which is variable in its capacitance value, and the two-terminal network  7  in parallel consisting of the parallel circuit comprising the fixed inductance  26  in parallel and the parallel capacitance element  13  which is variable in its capacitance value C(U)   b) symmetrical phase shifter with π structure formed from two two-terminal networks  7  in parallel as under a) and a two-terminal network  6  in series as under a).         
           [0010]      FIG. 2 : phase shifter according to the invention with T-structure with a variable-capacitance diode  15  as a series capacitance element  12  variable in its capacitance value in each of the two two-terminal networks  6  in series with a variable-capacitance diode  15  as a parallel capacitance element  13  which is variable in its capacitance value in the two-terminal network  7  in parallel, with supply of the variable-capacitance diodes  15  with the tuning voltage U by way of example for variation of the transmission phase Φ of the phase shifter. 
           [0011]      FIG. 3 : shows the phase shifter according to the invention with T-circuit as in  FIG. 2 , but without supply of the control voltage U. For adaptation of the range of variation of the series capacitance element  12  or the parallel capacitance element  13  to the range of variation of the variable-capacitance diode  15  concerned, a compensating series inductance  16  with the same inductance value LS as each other is connected in series. 
           [0012]      FIG. 4 :
   a) shows a variable-capacitance diode  15  or a varactor  15  with a self-inductance  18 , wherein a compensating series inductance  16  is connected in series in such a way that adaptation of the range of variation of the diode capacitance value to the range of variation of the capacitance element  12 ,  13  concerned is provided,   b) if the self-inductance  18  of the variable-capacitance diode is too high, adaptation of the range of variation is effected by a compensating parallel inductance  19  connected in parallel with the diode via a bypass capacitor  19 .     
           [0015]      FIG. 5 : phase shifter according to the invention with T-circuit as in  FIG. 3 , but with symmetrical, i.e. differential input  3  and differential output  4  of the two-port  2 . The centre line in dots and dashes shows the virtual earth  9  at which the circuit in  FIG. 3  is reflected. 
           [0016]      FIG. 6 :
   a) transmission loss in dB of a phase shifter which is optimised in its range of phase variation and its transmission loss according to  FIG. 3  with variable-capacitance diodes  15  of identical construction in all branches as a function of the tuned capacitance value C(k) of the capacitance elements  14  caused by the tuned voltage U   b) input reflection factor in dB of the phase shifter as under a) as a function of the capacitance value C(k) of the capacitance elements  14 .     
           [0019]      FIG. 7 : transmission phase Φ of the optimised phase shifter in  FIG. 6  as a function of the capacitance value C(k) of the variable-capacitance diodes  15 . 
           [0020]      FIG. 8 : phase shifter as a symmetrical two-port  2 , as in  FIG. 3 , made by microstrip conductor technology. 
       
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT 
       [0021]    The high-frequency phase shifter  1 , as shown as a symmetrical two-port  2  with its input  3  and its output  4 , consisting of 3 two-terminal networks  5  as a T-circuit in  FIG. 1   a  and as a π circuit in  FIG. 1   b , with suitable dimensioning has a particularly advantageously large range of variation of the transmission phase Φ with particularly low reflection loss. Here, all reactance elements are assumed to be low-loss. At the same time it must be particularly emphasised that the presence of the capacitance elements  12 ,  13 , which are electrically variable in their capacitance value both in the two-terminal networks  6  in series and in the two-terminal networks  7  in parallel, according to the invention affords the possibility of compensating to a particularly large extent the reflection at the ports caused by simultaneous variation of all capacitance values. Here it is of particular advantage that this aim is also achieved if all variable capacitance elements  12 ,  13  are the same in the two-terminal networks  6  in series and the two-terminal networks  7  in parallel, so that they can be embodied very advantageously by the same variable-capacitance diodes  15  or varactors. In this way capacitance elements which have low variance from each other and are formed on a semiconductor substrate can be used. The phase shifter already mentioned above and shown in DE 3802662A1 contains four high-frequency inductances. Its transmission phase is varied with only two variable-capacitance diodes. As a result a lower reflection loss is achieved, compared with the phase shifter of the present invention. As an added advantage over this state of the art, with the phase shifter according to the invention only three instead of four fixed inductances are used. By contrast with inductances, variable-capacitance diodes can be produced without high demand for surface area on a semiconductor substrate. In particular for producing a phase shifter in integrated technology, it is therefore substantially more economical to produce one inductance less and, in return, one variable-capacitance diode more on a substrate.
       The principal characteristics of the phase shifter  1  according to the invention are that each two-terminal network  7  in series consists of the series circuit comprising an identical fixed inductance  10  in series and an identical series capacitance element  12  which is electrically variable in its capacitance value   each two-terminal network  7  in parallel consists of the parallel circuit comprising an identical fixed inductance  26  in parallel and an identical parallel capacitance element  13  which is electrically variable in its capacitance value and   all capacitance elements in the symmetrical two-port  2  are varied in their capacitance value by the same tuning voltage U for variation of the transmission phase Φ of the two-port  2 .       
 
         [0025]    All two-terminal networks  6  in series of the phase shifters in  FIG. 1  are, with respect to their basic structure, designed as series resonant circuits tunable in their resonant frequency, with a fixed inductance  10  in series. Similarly all two-terminal networks  7  in parallel are, with respect to their basic structure, designed as parallel resonant circuits tunable in their resonant frequency, with a fixed inductance  26  in parallel, so that in each case there is a two-port  2  with the known basic structure of a resonant band-pass filter. 
         [0026]    By contrast with the resonant band-pass filter, of which the reflection loss is supposed to be low in a wider frequency range, for the phase shifter it is required that, in the vicinity of a discrete frequency f, it enables the transmission phase Φ variably, in each case with a high reflection loss. 
         [0027]    To accomplish this, special dimensioning of all reactance elements  8  of the two-port  2  is necessary for a phase shifter according to the present invention. 
         [0028]    To approach this, suitable standardisation for the reactance elements is carried out below for the elements in  FIG. 1 . 
         [0029]    For this, the fixed inductances  10  in series are denoted Ls and the fixed inductances  26  in parallel are denoted Lp. Furthermore, series and parallel capacitance elements  12 ,  13 ,  14  the same as each other, of which the capacitance value, depending on the tuning voltage U, has a value of C(U), are assumed. 
         [0030]    Standardisation lies in the following definitions: 
         [0000]    Let L0 be the geometric mean of Ls and Lp, so that: 
         [0000]        L 0 =√{square root over (Ls*Lp)}   
         [0031]    Let Ls be m times and L0 and Lp 1/m times as great as L0, so that: 
         [0000]    Ls=m*L0 and Lp=1/m*L0 
         [0032]    Let C0 be the capacitance value which together with the value L0 fulfils the resonance condition at the operating frequency f of the phase shifter, so that: 
         [0000]    
       
         
           
             
               C 
                
               
                   
               
                
               0 
             
             = 
             
               1 
               
                 
                   
                     ( 
                     
                       2 
                       * 
                       π 
                       * 
                       f 
                     
                     ) 
                   
                   2 
                 
                 * 
                 L 
                  
                 
                     
                 
                  
                 0 
               
             
           
         
       
     
         [0033]    Furthermore, let the resonant reactance referred to the reference characteristic impedance Z0 and formed from L0 and C0 be denoted X0, so that: 
         [0000]    
       
         
           
             
               X 
                
               
                   
               
                
               0 
             
             = 
             
               
                 
                   
                     L 
                      
                     
                         
                     
                      
                     0 
                   
                   
                     C 
                      
                     
                         
                     
                      
                     0 
                   
                 
               
               
                 Z 
                  
                 
                     
                 
                  
                 0 
               
             
           
         
       
     
         [0034]    For the effective capacitance value C(U) of the capacitance elements  14  or variable-capacitance diodes  15  which is varied by the tuning voltage U, the following shall hold true: 
         [0000]        C ( U )= k*C 0 
         [0000]    where k is varied by the tuning voltage U. 
         [0035]    Hence for the reference resonant frequency frs of the series resonant circuit we have: 
         [0000]    
       
         
           
             frs 
             = 
             
               1 
               
                 2 
                  
                 π 
                 * 
                 
                   
                     C 
                     * 
                     L 
                   
                 
                  
                 
                   
                     m 
                     * 
                     k 
                   
                 
               
             
           
         
       
     
         [0000]    and the reference resonant frequency frp of the parallel resonant circuit is: 
         [0000]    
       
         
           
             frp 
             = 
             
               
                 1 
                 
                   2 
                    
                   π 
                   * 
                   
                     
                       C 
                       * 
                       L 
                     
                   
                 
               
               * 
               
                 
                   m 
                   k 
                 
               
             
           
         
       
     
         [0036]    For the reactance value of the two-terminal network  6  in series, referred to Z0, we have as a function of X0, m and k: 
         [0000]        Xr =( m− 1 /k )* X 0 
         [0000]    and the susceptance of the two-terminal network  5  in parallel, referred to 1/Z0, is: 
         [0000]        Br =( k−m )/ X 0 
         [0037]    Depending on the quantities X0, m and the setting k for the complex transmission factor S21 of the T-structure circuit in  FIG. 1   a , the following relationship: 
         [0000]    
       
         
           
             
               S 
                
               
                   
               
                
               21 
             
             = 
             
               
                 
                   1 
                   
                     1 
                     + 
                     
                       j 
                        
                       
                           
                       
                        
                       Xr 
                     
                     + 
                     
                       
                         jBr 
                         2 
                       
                       * 
                       
                         
                           ( 
                           
                             1 
                             + 
                             jXr 
                           
                           ) 
                         
                         2 
                       
                     
                   
                 
                  
                 
                     
                 
                  
                 where 
                  
                 
                     
                 
                  
                 j 
               
               = 
               
                 
                   - 
                   1 
                 
               
             
           
         
       
     
         [0038]    By variation of the quantities X0 and m, the phase shifter can be optimised according to the respective requirements with respect to the extent of variation of the transmission phase Φ, taking the reflection loss into consideration. With the argument of S21, we have the transmission phase Φ. If the components are low-loss, the reflection factor is obtained from the amount of S21 with 
         [0000]        S 11=√{square root over (1 −S 21 2 )}
 
         [0039]    In  FIGS. 6 and 7  are shown by way of example the results which can be obtained with a phase shifter according to  FIGS. 2 and 3 . Here,  FIG. 6  shows the transmission phase Φ of a phase shifter optimised with respect to the extent of phase variation and reflection loss, as a function of the capacitance value C(k) of the variable-capacitance diodes  15 . The corresponding resulting reflection loss as well as the transmission loss are shown in  FIGS. 6   a  and  6   b . Particularly noteworthy here is the wide range of variation of the transmission phase Φ of nearly 200° which can be achieved in conjunction with a transmission loss of only 0.3 dB in the whole of the tuning range. 
         [0040]    In  FIG. 2  the phase shifter from  FIG. 1   a  is shown in more detail. The series capacitance elements  12  and the parallel capacitance element  13  are designed as variable-capacitance diodes  15  the same as each other. The tuning voltage U for varying the transmission phase Φ of the phase shifter is delivered to the node  20  and hence to the variable-capacitance diodes  15  on the one hand via the high-frequency choke coil  21  and on the other hand via the two-port earth  9 . The two variable-capacitance diodes  15  in the two two-terminal networks  6  in series receive the earth potential via the two high-frequency choke coils  21  which have high resistance at high frequency. The form of supply to the variable-capacitance diodes  15  which is shown is merely an example. It is in general determined rather by the technology which is used to produce the phase shifter. Therefore in the following figures, generation of the bias voltage of the variable-capacitance diodes  15  or varactors of the phase shifter is not shown. 
         [0041]    Often it is necessary to adapt the range of variation of the capacitance value CD(U) of a variable-capacitance diode  15  to the required range of variation of the effective capacitance C(U) of the capacitance elements  14  to achieve the required range of variation of the transmission phase Φ. In  FIG. 4   a  is shown a variable-capacitance diode  15  with its self-inductance  18 . Basically, this self-inductance is below the natural resonant frequency, increasing to the range of variation of the capacitance of the component. As a rule, however, it is frequently necessary to increase the range of variation of the capacitance C(U) of the capacitance elements  14  by series connection of a compensating series inductance  16  to the variable-capacitance diode  15 . If CD(U) is the capacitance of the inner variable-capacitance diode  15  and LS is the effective series inductance, consisting of the self-inductance  18  of the component and the compensating series inductance  16  together, then the effective capacitance value C(U) of the capacitance element  14  is obtained as: 
         [0000]    
       
         
           
             
               C 
                
               
                 ( 
                 U 
                 ) 
               
             
             = 
             
               
                 CD 
                  
                 
                   ( 
                   U 
                   ) 
                 
               
               * 
               
                 1 
                 
                   1 
                   - 
                   
                     
                       
                         ( 
                         
                           2 
                            
                           π 
                            
                           
                               
                           
                            
                           f 
                         
                         ) 
                       
                       2 
                     
                     * 
                     LS 
                     * 
                     
                       CD 
                        
                       
                         ( 
                         U 
                         ) 
                       
                     
                   
                 
               
             
           
         
       
     
         [0042]    If the self-inductance  18  of the variable-capacitance diode is too high, it may be necessary to narrow the range of variation of the effective capacitance value C(U) by parallel connection of a compensating parallel inductance  18  via a bypass capacitor  19 , as shown in  FIG. 4   b.    
         [0043]    In  FIG. 3  are shown the capacitance elements  4  which are the same as each other, each with a compensating series inductance  16 . To produce the phase shifter, in the two-terminal networks  6  in series the fixed inductance  10  in series and the compensating inductance  16  in series are in each case appropriately combined into one component. 
         [0044]    For frequencies in the gigahertz frequency range, the high-frequency phase shifter  1  may advantageously be constructed in microstrip conductor technology as a symmetrical two-port  2 , as shown in  FIG. 8 . Here, the fixed inductances  10  in series and the compensating inductances  16  in series and fixed inductance  26  in parallel are structured as short electrical wires. 
         [0045]    In  FIG. 5  the high-frequency phase shifter  1  is designed as a differential phase shifter with input and output symmetrical in relation to the two-port earth  9 , in such a way that the symmetrical two-port  4  is reflected at the two-port earth  9 , so that in each case the two fixed inductances  26  in parallel in the two-terminal networks  7  in parallel are combined into a common fixed inductance  27  in parallel, and also the series-connected variable-capacitance diodes  15  are combined into a parallel capacitance element  13 . Symmetrical arrangements of this kind are particularly suitable for the technology of integrated circuits. The reactances and in particular the variable-capacitance diodes  15  are manufactured on a semiconductor substrate in the interests of low variance of the capacitance values from each other.