Abstract:
The apparatus for polarity-inversion-protected supplying of an electronic component with an intermediate voltage from a supply voltage has a first transistor connected between a terminal for the supply voltage and the electric component, so that a path between a source terminal and a sink terminal is flown through by a supply current of the electric component, a potential-to-current conversion device formed so that it generates a rising response current at a rising voltage drop across the path between source terminal and sink terminal of the first transistor, and a coupler formed to provide a follow current proportional to the response current at an output of the coupler depending on the response current, wherein the output of the coupler is coupled to a control terminal of the first transistor, wherein the potential-to-current conversion means has a second transistor providing the response current, and wherein the potential-to-current conversion device further has a potential shifter.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS  
       [0001]     This application claims priority from German Patent Application No. DE 10 2005 040 072.8, which was filed on Aug. 24, 2005, and is incorporated herein by reference in its entirety.  
       TECHNICAL FIELD  
       [0002]     The present invention relates to an apparatus for polarity-inversion-protected supplying of an electronic component with an intermediate voltage from a supply voltage, and particularly to an apparatus for polarity-inversion-protected supplying of semiconducting switching elements and arrangements of semiconducting switching elements, such as chips.  
       BACKGROUND  
       [0003]     High demands are made on the supply of electric components with electric energy, and here particularly on the supply of semiconducting chips. An important demand is to perform the energy supply in polarity-inversion-protected manner. Within the scope of the present application, by “polarity-inversion-protected supply” of an electric component or of a chip, a supply with electric voltage is understood, in which the input resistance at a terminal designed for a positive voltage for example becomes sufficiently high-ohmic when applying a negative voltage, so that the resulting current flow does not lead to thermal overload. Moreover, all elements also need to have the required voltage strength.  
         [0004]     A first possible embodiment of a polarity-inversion-protected supply of an electric component is to connect a diode or a transistor connected as a diode between an external supply voltage Vsup,ext and a regulation circuit providing an internal regulated supply voltage Vsup,int, so that the diode is operated in pass direction when a positive supply voltage Vsup,ext is applied, and in blocking direction in the case of a negative external supply voltage Vsup,ext. Here, within the scope of the present application, in the case of transistors, an emitter terminal of a bipolar transistor or a source terminal of a field-effect transistor is understood by a source terminal, a collector terminal of a bipolar transistor and a drain terminal of a field-effect transistor by a sink terminal, as well as a base terminal of a bipolar transistor and a control terminal or gate terminal of a field-effect transistor by a control terminal.  
         [0005]      FIG. 2  shows such a circuit of a possible realization of a polarity-inversion-protected supply of an electric component.  FIG. 2  shows a series connection of a pnp bipolar transistor  800  and an npn bipolar transistor  810 , which are connected to the external supply voltage Vsup,ext with an emitter terminal of the pnp bipolar transistor  800  and to a regulated internal supply voltage Vsup,int with an emitter terminal of the npn bipolar transistor  810 . A collector terminal of the pnp bipolar transistor  800  is connected to a collector terminal of the npn bipolar transistor  810  and a base terminal of the pnp bipolar transistor  800 .  
         [0006]     A base terminal of the npn bipolar transistor  810  is connected to a regulating means not included in  FIG. 2 . The regulating means not shown in  FIG. 2  regulates the internal, regulated supply voltage Vsup,int from a voltage VnetX present at the collector terminal of the npn bipolar transistor  810 . The regulating means is not shown in  FIG. 2 , since it is not subject matter of the present application, but rather known.  
         [0007]     The npn bipolar transistor  810 , via which the actual regulation of the internal regulated supply voltage Vsup,int is done, is, however, not polarity-inversion-protected with reference to a positive external supply voltage Vsup,ext. This is due to the fact that, especially with integrated circuits constructed on a semiconductor substrate doped with holes (p substrate), the diode formed between collector and substrate is operated in pass direction and destroyed due to the high voltage present and the current flow resulting therefrom, when applying a great-magnitude negative voltage (e.g. −20 V) to the area doped with electrons, which forms the collector of the npn bipolar transistor  810  (n collector).  
         [0008]     In other words, if the intermediate voltage Vsup,ext (e.g. 3 V . . . 30 V) is applied to the IC (integrated circuit), the IC generates an internal, regulated voltage Vsup,int (e.g. 2.5 V), by suitably regulating the base of the npn bipolar transistor  810  (not shown, since prior art). Here, the npn bipolar transistor  810  functions as a series regulator (pass transistor). The collector of the npn bipolar transistor  810  is, however, not polarity-inversion-protected: i.e., when applying a voltage negative as opposed to the p substrate lying at ground potential or at a reference potential (e.g. −20 V), the collector-substrate diode opens, draws a lot of current and is destroyed.  
         [0009]     A possible polarity inversion protection is to connect a diode or a pnp bipolar transistor  800  connected as a diode, as  FIG. 2  shows, between the external supply potential Vsup,ext and the collector terminal of the npn bipolar transistor  810 . A connection of a pnp bipolar transistor, as shown by  FIG. 2  for the pnp bipolar transistor  800 , in which the base terminal and the collector terminal are connected or shorted by a low-ohmic line, is also referred to as a transistor diode. In case of a polarity inversion, a voltage negative as opposed to the reference potential is present at the area doped with holes of the semiconducting substrate, which forms the emitter of the pnp bipolar transistor  800 , so that the pnp bipolar transistor  800  blocks.  
         [0010]     In other words, the above described opening of the collector-substrate diode, drawing a large current, and the destruction of the npn bipolar transistor  810 , which will also be referred to as regulating transistor in the following, is prevented in the prior art by laying a pnp transistor  800  connected as a diode between the collector of the npn bipolar transistor and the external supply pin. Thus, the p emitter of the pnp bipolar transistor  800 , which blocks at a negative potential as opposed to the substrate (negative voltage at p), is located at an external supply pin.  
         [0011]     The disadvantage of this solution is that the transistor diode  800  requires at least a potential difference of a flow voltage between its emitter terminal and its collector terminal, so as to become conducting. Within the scope of the present application, by a flow voltage, a potential difference between an anode terminal and a cathode terminal of a diode or between a base and an emitter terminal of a bipolar transistor is understood, which leads to a current flow through the diode or to a collector current through the bipolar transistor of 1 mA, wherein the diode and the bipolar transistor are operated in pass direction. A flow voltage thus corresponds to about the voltage at which a bend occurs in case of a diode characteristic curve or the transmission characteristic curve of a bipolar transistor.  
         [0012]     In case of a diode or a transistor diode based on silicon, a flow voltage thus typically ranges from about 500 mV to 800 mV. Since a voltage between 20 mV and 250 mV, which is also referred to as saturation voltage and corresponds to the output characteristic curve field of a typical bipolar transistor of the collector-emitter voltage at which the output characteristic curves have a bend, also typically drops in a bipolar transistor across the path between a collector terminal and an emitter terminal at a collector current of 1 mA, the circuit in  FIG. 2  requires an external supply voltage Vsup,ext exceeding at least the internal, regulated supply voltage Vsup,int by the sum of a flow voltage and a saturation voltage. This means that the external supply voltage Vsup,ext has to be greater than 3.5 V, for example, at an internal, regulated supply voltage Vsup,int of about 2.5 V.  
         [0013]     In other words, the disadvantage of the circuit shown in  FIG. 2  is that the diode  800  requires at least a flow voltage so as to become conducting. This means, when for example Vsup,int is regulated to 2.5 V, the circuit only functions for external supply voltages Vsup,ext of about 3.5 V and above at low temperatures, because then the base-emitter voltage of the bipolar transistor  800  becomes about 0.9 V and the npn bipolar transistor  810  still needs at least 0.1 V saturation voltage so as to be able to work.  
         [0014]     A further procedure to realize a possible polarity inversion protection is the so-called “low-drop” technology. Here, the regulation of the internal, regulated supply voltage Vsup,int from the external supply voltage Vsup,ext is no longer done via an npn bipolar transistor, as it is shown in  FIG. 2 , but a pnp bipolar transistor is rather used also for the regulation of the internal supply voltage Vsup,int. In other words, the npn emitter follower may of course be omitted and constitute a series regulator solely with a pnp bipolar transistor. This procedure solves the problem of the flow voltage occurring in addition, as it occurs in an additional series-connected diode or transistor diode. In this case, the circuit functions as long as the external supply voltage Vsup,ext exceeds the internal, regulated supply voltage Vsup,int by at least the saturation voltage, i.e. typically between 20 and 250 mV.  
         [0015]     In other words, this procedure solves the problem of the drop voltage: The circuit functions as long as the external supply voltage Vsup,ext is greater than the sum of the regulated, internal supply voltage Vsup,int and about 100 mV. The disadvantage of this solution is that the internal, regulated supply voltage Vsup,int now lies at a collector of a pnp bipolar transistor, which is substantially more high-ohmic than the emitter of the npn bipolar transistor  810  in  FIG. 2 . The reason for the higher internal resistance of the collector than that of the emitter is that the former constitutes a current source, but the latter a voltage source, and lies in the slope of the Ice-Uce output characteristic curve (output characteristic curve Ice versus Uce) as well as the Ie-Ube input characteristic curve (input characteristic curve Ie versus Ube). Thereby, it becomes much more difficult to keep the locked loop of the internal, regulated supply voltage Vsup,int stable, particularly when the capacitive load at Vsup,int has not exactly been determined yet. So as to guarantee the regulation of the internal, regulated supply voltage Vsup,int in this case, the internal, regulated supply voltage Vsup,int has to be loaded to the reference potential or also to ground via a stabilizing load or also a shunt path. The current flowing off through the stabilizing load is used for stabilizing the regulating circuit and thus reduces the current efficiency of such a current regulation, wherein the current efficiency is defined as the ratio of the current the regulation gives off to the actual load connected thereto divided by the overall current consumption, i.e. including the own current consumption of the regulation.  
         [0016]     In other words, a relatively large current (as compared with the overall current need of the circuit) has to be sunk from Vsup,int via a shunt path or stabilizing load to ground, in order to keep the locked loop stable. This current is used for achieving the stability and wastes and reduces the efficiency of such a voltage regulation: For example, if Vsup,int is loaded with 5 mA, about 1.7 mA additional current is needed in order to keep the low-drop regulator stable. Its efficiency thus is about 75%, whereas the efficiency of the emitter follower shown in  FIG. 2  is far higher at about 90%.  
         [0017]      FIG. 3  shows an example for such a low-drop circuit. A pnp bipolar transistor  820  is connected to the external supply voltage Vsup,ext with an emitter terminal and to the regulated, internal supply voltage Vsup,int with a collector terminal. A base terminal of the pnp bipolar transistor  820  is connected to a regulating unit not included in  FIG. 3  since it is known and does not represent the subject matter of the present application. Moreover,  FIG. 3  shows an NMOS transistor  830  connected between the regulated, internal supply voltage Vsup,int and a reference potential.  
         [0018]     The pnp bipolar transistor  820  here serves for the regulation of the internal supply voltage Vsup,int. The NMOS transistor  830  serves for loading the regulated, internal supply voltage Vsup,int and thus represents the stability load explained further above. Alternatively, instead of the NMOS transistor  830 , also a PMOS transistor may be used as stability load. The advantage of this solution is that again a low-ohmic source terminal lies at the terminal for Vsup,int, and not a high-ohmic drain terminal.  
       SUMMARY  
       [0019]     An improved concept for polarity-inversion-protected supply of an electronic component with an intermediate voltage, wherein the supply is to be guaranteed also at only slightly higher external supply voltage and is to have high current efficiency, provides an apparatus for polarity-inversion-protected supplying of an electronic component with an intermediate voltage from a supply voltage, having: a first transistor connected between a terminal for the supply voltage and the electric component, so that a path between a source terminal and a sink terminal of the first transistor is flown through by a supply current of the electric component; a potential-to-current converter formed so that it generates a rising response current at a rising voltage drop across the path between source terminal and sink terminal of the first transistor; and a coupler formed to provide a follow current proportional to the response current at an output of the coupler depending on the response current, wherein the output of the coupler is coupled to a control terminal of the first transistor; wherein the potential-to-current converter has a second transistor providing the response current; and wherein the potential-to-current converter further has a potential shifter coupled to the sink terminal of the first transistor and a control terminal of the second transistor, and formed to generate a potential dependent on a potential at the sink terminal of the first transistor at the control terminal of the second transistor. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0020]     These and other objects and features of the present invention will become clear from the following description taken in conjunction with the accompanying drawings, in which:  
         [0021]      FIG. 1  is a circuit diagram of a polarity-inversion-protected current supply circuit according to an embodiment;  
         [0022]      FIG. 2  is a circuit diagram of a first possible embodiment of a polarity-inversion-protected current supply circuit; and  
         [0023]      FIG. 3  is a circuit diagram of a second possible embodiment of a polarity-inversion-protected current supply circuit in “low-drop” technology. 
     
    
     DETAILED DESCRIPTION  
       [0024]     Different embodiments are based on the finding that polarity inversion protection for the supply of an electronic component with an intermediate voltage from a supply voltage can be realized by connecting a transistor with a source terminal and a sink terminal between the supply voltage and the electronic component so that a supply current of the electronic component flows through the source terminal and the sink terminal of the transistor. Here, a control terminal of the transistor may be supplied with a voltage or a current by a regulating circuit, so that the voltage drop across the path between the source terminal and the sink terminal of the transistor is smaller than the voltage drop that would occur in the connection of a transistor as a diode in pass direction. Moreover, it is possible to design the regulating circuit coupled to the control terminal of the transistor so that it has little current consumption of its own as compared with the current need of the electronic component.  
         [0025]     A preferred embodiment comprises such a potential-to-current conversion means, which makes a current available depending on the potential at the sink terminal of the transistor of a coupling means. A follow current, on the basis of which the second transistor is driven, is generated by the coupling means at an output of the coupling means.  
         [0026]     With reference to  FIG. 1 , an embodiment of a circuit for polarity-inversion-protected supply of an electronic component with an intermediate voltage from a supply voltage will be described in the following.  
         [0027]      FIG. 1  shows a circuit diagram of an embodiment of a polarity-inversion-protected supply circuit  100 . A first transistor  110 , embodied as a pnp bipolar transistor, is coupled to a source terminal or emitter terminal at an external supply voltage Vsup,ext. Within the scope of the present application, in the case of transistors, here an emitter terminal of a bipolar transistor or a source terminal of a field-effect transistor is understood by a source terminal, a collector terminal of a bipolar transistor and a drain terminal of a field-effect transistor by a sink terminal, as well as a base terminal of a bipolar transistor and a control terminal or gate terminal of a field-effect transistor by a control terminal. A sink terminal or collector terminal of the first transistor  110 , which also represents the output of the polarity-inversion-protected supply circuit  100 , is connected to a collector terminal of a regulating transistor  120  embodied as an npn bipolar transistor and also referred to as series regulator. An emitter terminal of the regulating transistor or series regulator  120  is connected to the internal, regulated supply voltage Vsup,int. A control terminal or base terminal of the regulating transistor  120  is connected to a regulating circuit not illustrated in  FIG. 1 , since a possible solution is known and not the subject matter of the present application. A potential VnetX is present at the sink terminal of the first transistor  110 .  
         [0028]     Apart from the first transistor  110 , the supply circuit  100  also includes a potential-to-current conversion means  140  also referred to as voltage-current converter or V-I converter, and a coupling means  150 . An input of the potential-to-current conversion means  140  is coupled to the external supply voltage Vsup,ext. An output of the potential-to-current conversion means  140  is connected to an input of the coupling means  150 . The V-I converter  140  includes a first resistor  130 , which comprises a first terminal representing the input of the V-I converter  140 , a second transistor  160  embodied as a pnp bipolar transistor, and a potential shifting means  170 . A second terminal of the first resistor  130  is connected to an emitter terminal of the second transistor  160  of the V-I converter  140 , which also represents the input of the potential-to-current conversion means  140 . A collector terminal of the second transistor  160 , which also represents the output of the potential-to-current conversion means  140 , is connected to an input of the coupling means  150 . A control terminal of the second transistor  160  is connected to an output of the potential shifting means  170 . The potential shifting means  170  includes a third transistor  180  and a second resistor  190 . The third transistor  180  is embodied as a pnp bipolar transistor in the present embodiment. The second resistor  190  is connected between a sink terminal or collector terminal of the third transistor  180  and a reference potential. A source terminal or emitter terminal of the third transistor  180 , which represents an input of the potential shifting means  170  and a further input of the V-I converter  140 , is connected to the sink terminal or collector terminal of the first transistor  110 . Moreover, the collector terminal of the third transistor  180  is coupled to a control terminal or a base terminal of the third transistor  180  via a low-ohmic, conducting connection, which will also be referred to as shorted in the further course of the present application, so that the third transistor  180  is operated as a diode. Moreover, the two control terminals or base terminals of the second transistor  160  and the third transistor  180  are connected to each other. The second transistor  160  and the third transistor  180  thus are of the same type. Generally speaking, this means that both transistors are designed as pnp bipolar transistors, as npn bipolar transistors, as NMOS transistors, or as PMOS transistors.  
         [0029]     An output of the coupling means  150  is connected to the control input of the first transistor  110  on the one hand and to a first terminal of a drive circuit  200  on the other hand. A second terminal of the drive circuit  200  is coupled to the external supply voltage Vsup,ext with a further terminal. The drive circuit  200  includes a series connection of a third resistor  210  and a fourth transistor  220 , which is embodied as a pnp bipolar transistor, wherein an emitter terminal of the fourth transistor  220  is connected to the third resistor  210  and a collector terminal of the fourth transistor  220  to the first terminal of the drive circuit  200  on the one hand, and a control terminal or base terminal of the fourth transistor  220  on the other hand, so that the fourth transistor  220  is connected as a diode.  
         [0030]     The coupling means  150  includes a fifth NMOS transistor  230  and a sixth NMOS transistor  240 , wherein a control terminal or gate terminal of the fifth NMOS transistor  230  is connected to a control terminal of the sixth NMOS transistor  240  as well as to a sink terminal or drain terminal of the fifth NMOS transistor  230 , which also represents the input of the coupling means  150 . A sink terminal or drain terminal of the sixth NMOS transistor  240  represents the output of the coupling means  150 . A source terminal of the fifth NMOS transistor  230  and a source terminal of the sixth NMOS transistor  240  are connected to the ground potential (GND). The fifth NMOS transistor  230  and the sixth NMOS transistor  240  thus form a current mirror circuit.  
         [0031]     In the following, the functioning of the above-described circuitry, which is a current-efficient, polarity-inversion-protected low-drop regulator, will be explained in greater detail. The embodiment shown in  FIG. 1  solves the problem of the voltage drop, also referred to as drop, across the first transistor  110  acting as a diode as a result of the connection of the pnp bipolar transistor  800  in possible circuits, as this is illustrated in  FIG. 2 , and the problem, which occurs in possible circuits, of the current sunk for stabilization of the regulating circuit, which is also referred to as shunt current, as it has been explained in connection with the circuit illustrated in  FIG. 3 , which shows one possible solution. These two problems are solved by regulating the potential at the output of the polarity-inversion-protected supply circuit  100 , and thus at the collector terminal of the first transistor  110 , to a value VnetX, which typically lies between 20 mV and 500 mV, preferably 50 mV to 200 mV, below the external supply voltage Vsup,ext. The voltage or potential difference VnetX as opposed to the reference potential is also referred to as intermediate voltage, and the accompanying potential also as intermediate potential, whereas Vsup,ext−VnetX is the collector-emitter voltage of the first transistor  110 . In other words, the problem of the drop of the possible realization of a polarity-inversion-protected voltage supply in  FIG. 2  and of the shunt current of the possible realization of a polarity-inversion-protected voltage supply in  FIG. 3  can be solved by a regulating circuit regulating the potential at a node at the collector terminal of the first transistor  110  to a value lying about  100  mV below the external supply voltage Vsup,ext.  
         [0032]     To this end, the potential VnetX present at the collector terminal of the first transistor  110  is made available to the potential shifting means  170  as an input signal. Due to the potential difference present at the series connection of the third transistor  130  connected as a diode and the second resistor  190 , a current flows off via the collector terminal and the emitter terminal of the third transistor  180  and the second resistor  190 . Due to the series connection of the third transistor  180  and the second resistor  190 , a potential differing from the potential VnetX present at the collector terminal of the first transistor  110  by about a flow voltage of a pn junction is present at the output of the potential shifting means  170 , i.e. the base terminal of the third transistor  180 . Typically, the potential difference between the base input of the second transistor  160  and the collector terminal of the first transistor  110 , i.e. the potential at the base terminal of the second transistor  160  and the potential VnetX, deviates from a flow voltage of a pn junction by less than 50% of a flow voltage of a pn junction, and preferably by no more than  20  % of a flow voltage of a pn junction. Within the scope of the present application, by a flow voltage, a potential difference between an anode and a cathode terminal of a diode or between a base terminal and an emitter terminal of a bipolar transistor is understood, which leads to a current flow through the diode or a collector current through the bipolar transistor of 1 mA, wherein the diode and the bipolar transistor are operated in pass direction. A flow voltage thus corresponds to about the voltage at which a bend occurs in case of a diode characteristic curve or the transmission characteristic curve of a bipolar transistor. In case of a diode or a transistor, which is based on silicon, a flow voltage thus typically ranges from about 500 mV to 800 mV. As a result, a potential difference between the collector terminal of the first transistor  110  and the base terminal of the second transistor  160 , which is connected to the output of the potential shifting means  170 , of typically 0.1 to 1.0 V, and preferably from 0.6 to 0.8 V, results this way. In other words, the potential at the base terminal of the second transistor  160  preferably ranges from about 0.6 V to 0.8 V below the potential at the collector terminal of the first transistor  110 .  
         [0033]     The first resistor  130  and the second resistor  190 , as well as the second transistor  160  and the third transistor  180 , are designed so that, in a rest state or equilibrium state of the locked loop or regulating circuit, a voltage deviating by no more than typically 50%, preferably 20%, from the voltage drop between the emitter terminal and the collector terminal of the first transistor  110  drops for an external supply voltage Vsup,ext, which lies above the intermediate potential VnetX by less than 1.5 V, due to an emitter current of the second transistor  160 . Within the scope of the present application, by a rest state or equilibrium state of the regulating circuit or locked loop, a distribution of currents and potentials or voltages is understood, which are temporally constant in an ideal system. In this connection, within the scope of the present application, by an ideal system, a circuit is understood, the components of which comprise no random voltage fluctuations or current fluctuations generally referred to as noise.  
         [0034]     In other words, for the regulation of the potential VnetX to a value of about Vsup,ext −100 mV, a current is drained to the reference potential (GND) or to ground via the third transistor  180  and second resistor  190 , which current thus generates a potential generating the same voltage across the first resistor  130  in a rest state of the locked loop as is present across the collector-emitter path of the first transistor  110  at the base terminal of the second transistor  160 . In still other words, in the rest state of the locked loop, a voltage that is ideally equal to the voltage across the collector-emitter path of the first transistor  110  is generated across the first resistor  130 .  
         [0035]     In still other words, the first resistor  130  causes an emitter potential at the emitter terminal of the second transistor  160 , which corresponds to the intermediate potential VnetX, and thus determines a dependence of the current given off to the collector terminal of the second transistor  160 , and of the intermediate potential VnetX. A resistance R 130  of the first resistor  130  thus determines, in a first approximation, a linear relationship between a voltage V and a current I flowing at the output of the V-I converter  140 , which meets the dependence I=V/R 130  in a first approximation.  
         [0036]     In the present embodiment, the coupling means  150  is formed so that it provides, at its output, i.e. at the sink terminal or drain terminal of the sixth NMOS transistor  240 , a follow current deviating from a collector current of the second transistor  160  and/or the response current of the V-I converter  140  at the input of the coupling means  150 , i.e. at the sink terminal or drain terminal of the fifth NMOS transistor  230 , by typically no more than 30%, preferably by no more than 10%, of the response current.  
         [0037]     By the coupling means  150  and/or by the current mirror circuit including the fifth NMOS transistor  230  and the sixth NMOS transistor  240 , a follow current proportional to the collector current of the second transistor  160  is provided at the output of the coupling means  150 . The drive circuit  200  connected to the external supply voltage Vsup,ext at its second terminal and including the third resistor  210  and the fourth resistor  220  and connected to both the base terminal of the first transistor  110  and the output of the coupling means  150  with its first terminal has a non-linear current-voltage characteristic curve. Due to the series connection of the third resistor  210  and the fourth transistor  220  connected as a diode, the drive circuit  200  provides, at its first terminal, a potential which is smaller as opposed to the external supply voltage Vsup,ext by a potential difference dependent on a current flowing through the drive circuit  200  and given by about a sum of an ohmic voltage drop across the third resistor  210  and a flow voltage of a pn junction. Thus, depending on the follow current, a potential is provided at the base terminal of the first transistor  110  (or a base-emitter voltage of the first transistor  110  is provided), so that a voltage of typically 20 mV to 500 mV, preferably from 50 to 200 mV, drops in the equilibrium state of the regulation or the regulation circuit across the path between the emitter terminal and the collector terminal of the first transistor  110 .  
         [0038]     In other words, the potential present at the control input of the second transistor  160  generates a current flow copied, via the NMOS transistors  230  and  240 , to a current path including the third resistor  210  and the fourth transistor  220  connected as a diode. Part of the current supplied from the sixth NMOS transistor  240  is further fed to the first transistor  110  as base current.  
         [0039]     If the potential at the sink terminal (collector terminal) of the first transistor  110 , i.e. the intermediate voltage VnetX, drops below the value of the intermediate voltage in the equilibrium state, a lower current flows to the reference potential via the input of the potential shifting means  170 , the collector-emitter path of the third transistor  180 , and the second resistor  190 . As a result, the potential present at the base terminal of the transistor  160  decreases, so that the base-emitter voltage of the second transistor  160  rises, which in turn results in the fact that the collector current through the second transistor  160  increases. Due to the current mirror circuit including the two NMOS transistors  230  and  240 , a higher follow current also flows through the drive circuit  200 , so that a potential reduced as opposed to the equilibrium state is present at the base terminal of the first transistor  110 . Due to the base-emitter voltage thereby increasing at the first transistor  110 , the potential at the collector terminal of the first transistor  110  rises again, which closes the locked loop.  
         [0040]     In other words, if the potential VnetX at the collector terminal of the first transistor  110  is too small in a non-equilibrium state of the regulating circuit, i.e. smaller than in the equilibrium state of the regulating circuit, the current flow through the third resistor  210  increases and opens the first transistor  110 , so that an operating point, in which the first transistor is, shifts such that a collector-emitter voltage of the first transistor  110  decreases. Correspondingly, the collector terminal of the first transistor is pulled up or a greater current fed into the node at the collector terminal of the first transistor  110 , which counteracts the voltage drop, which is increased as opposed to the equilibrium state, across the collector-emitter path of the first transistor  110 .  
         [0041]     In the design of the supply circuit  100 , i.e. particularly in the design of the resistors  130 ,  190 , and  210  as well as the transistors  110 ,  160 ,  180 ,  220 ,  230 , and  240 , achieving a difference between the potential of the external supply voltage Vsup,ext and the potential of the intermediate voltage VnetX of typically 20 mV to 500 mV and preferably from 50 mV to 200 mV is only necessary at values of the external supply voltage Vsup,ext typically lying above the internal, regulated supply voltage Vsup,int of the electronic component by less than 1.5 V and preferably by less than 300 mV. With an external supply voltage exceeding the value of the internal, regulated supply voltage Vsup,int by more than the above-mentioned voltages, a greater voltage drop across the path between the collector terminal and the emitter terminal of the first transistor  110  may also occur, without the basic functionality of the electronic component being at risk. In other words, this regulation only has to be exact at a minimum external supply voltage Vsup,ext. At greater external supply voltages Vsup,ext, the regulation is also allowed to permit greater differences between the external supply voltage Vsup,ext and the intermediate voltage VnetX of more than 100 mV, because in this case also a greater voltage is available. Rather, it is even of advantage if the voltage difference Vsup,ext−Vsup,int is distributed to the collector-emitter path of the first transistor  110  and of the regulating transistor  120  as uniformly as possible at great external supply voltages Vsup,ext, because the two transistors are then loaded minimally, and the voltage strength of the circuit thus increases.  
         [0042]     A further advantage the supply circuit  100  shown in  FIG. 1  has is that it comprises no extremely high-ohmic point, and that its output, which matches the collector terminal of the transistor  110 , is not highly loaded capacitively, so that its frequency stability generally does not present a problem. In individual cases, it may be necessary to connect a frequency compensation capacitance between the base terminal and the collector terminal of the first transistor  110 .  
         [0043]     Moreover, the circuit may be designed so that the additional current need due to the transistors  160 ,  180 , and  220  may be limited to typically less than 0.5 mA and preferably to less than 0.2 mA, so that the efficiency or the current efficiency of the voltage regulation is not degraded significantly. The additional space requirement on a chip also is not substantial as compared to a circuit according to  FIG. 2 , which shows one possible solution, since the circuitry only comprises three additional pnp bipolar transistors, three resistors, and two NMOS transistors in comparison with the circuit shown in  FIG. 2 , the space requirement of the two NMOS transistors being negligible in general.  
         [0044]     In other words, the polarity inversion protection is still realized by a pnp transistor, but it is not connected as a diode, as the possible solutions in  FIG. 2  and  FIG. 3  show, or used directly for the regulation of the low-drop voltage regulator, but a stand-alone locked loop is constructed therewith, which regulates the potential at the collector of the npn emitter follower representing the series regulator to about Vsup,ext−100 mV. Technologically speaking, this can be realized more easily than usual low-drop regulators and consumes less current and space.  
         [0045]     In other words, a regulated supply voltage Vsup,int, which is still guaranteed at a small external supply voltage Vsup,ext=Vsup,int+0.1 V, will be provided at the chip or on a chip. Here, the current efficiency of this regulation is about 90%. Furthermore, the circuit is polarity-inversion-protected, i.e. it is not destroyed even by applying negative voltages of about −20 V to a supply voltage pin. The embodiment illustrated in  FIG. 1  thus represents a current supply with a polarity inversion protection with reduced drop and high efficiency.  
         [0046]     Although the transistors  110 ,  120 ,  160 ,  180 , and  220  are embodied as bipolar transistors in the preferred embodiment, these may also be embodied as field-effect transistors. Likewise, the two field-effect transistors  230  and  240 , embodied as NMOS transistors in the preferred embodiment, may also be embodied as bipolar transistors. In particular, the regulating transistor or series regulator  120  may be embodied as a depletion NMOS transistor or self-conducting NMOS transistor, because driving this transistor can be done with a voltage lying below the internal, regulated supply voltage Vsup,int. Likewise, a circuit complementary to the circuitry shown in  FIG. 1  may be used for polarity inversion protection at a negative external supply voltage, in which the npn bipolar transistors occurring may be exchanged for pnp bipolar transistors, the NMOS transistors occurring for PMOS transistors, and vice versa.  
         [0047]     Moreover, a frequency compensation means, for example a Miller frequency compensation capacitance, may be connected between the control terminal or gate terminal and the sink terminal or drain terminal of the sixth NMOS transistor  240 , in order to enhance the frequency stability. Likewise, a further frequency compensation means, for example a Miller frequency compensation capacitance, may be connected between the base terminal or the control terminal and the collector terminal or the sink terminal of the first transistor  110 , in order to enhance the frequency stability.  
         [0048]     The coupling means  150  embodied as current mirror circuit in the embodiment explained and including the NMOS transistors  230  and  240  may be replaced by a different coupling means, preferably a current mirror circuit with or without scaling of the current flow. For example, scale factors or mirror ratios of 1:2 or also 2:1 are absolutely possible. In principle, however, also a coupling means  150  with a scale factor deviating from these scale factors or mirror ratios, which are only to be seen as exemplary, may be employed. But a mirror ratio of about 1:1 is often more favorable. One possible embodiment for this is, for example, a Wilson current mirror, which can guarantee smaller deviation of the follow current from the response current as compared to the above-described coupling means  150 .  
         [0049]     The drive circuit  200  and the potential shifting means  170  explained in the above embodiment and both constructed from a resistor  210  and  190  and a transistor  220  and  180  connected as a diode in the preferred embodiment, respectively, may be constructed by a different arrangement of circuit elements, such as ohmic resistors, diodes, or other electronic components.  
         [0050]     While this invention has been described in terms of several preferred embodiments, there are alterations, permutations, and equivalents which fall within the scope of this invention. It should also be noted that there are many alternative ways of implementing the methods and compositions of the present invention. It is therefore intended that the following appended claims be interpreted as including all such alterations, permutations, and equivalents as fall within the true spirit and scope of the present invention.