Abstract:
In a current mode controlled switching power supply, current through the inductor is sensed to determine when to turn off or on the switching transistors. The inductor current has a higher frequency AC component and a lower frequency DC component. The AC current feedback path, sensing the ramping ripple current, is separate from the DC current path, sensing the lower frequency average current. Separating the current sensing paths allows the signal to noise ratio of the AC sense signal to be increased and allows the switching noise to be filtered from the DC sense signal. The gain of the DC sense signal is adjusted so that the DC sense signal has the proper proportion to the AC sense signal. The AC sense signal and the DC sense signal are combined by a summing circuit. The composite sense signal is applied to a PWM comparator to control the duty cycle of the switch.

Description:
FIELD OF THE INVENTION 
       [0001]    This invention relates to DC/DC converters and, in particular, to sensing the current in a current mode controlled switching power supply. 
       BACKGROUND 
       [0002]      FIG. 1  illustrates one type of prior art current mode DC/DC switching power supply, also known as a current mode DC/DC converter. Many other converter configurations can also benefit from the present invention. The type of converter shown in  FIG. 1  is a peak current mode converter. 
         [0003]    The operation of the converter is conventional and is as follows. 
         [0004]    A clock (Clk) signal is applied to the set input of an RS flip flop  20 . 
         [0005]    The setting of the RS flip flop  20  generates a high signal at its Q output. A logic circuit  24 , in response, turns transistor switch  26  on and turns the synchronous rectifier switch  28  off Both switches may be MOSFETs or other transistors. A diode may replace the synchronous rectifier switch  28 . The logic circuit  24  ensures that there is no cross-conduction of switches  26  and  28 . The input voltage Vin applied to an inductor L 1  through the switch  26  causes a ramping current to flow through the inductor L 1 , and this current flows through a low value sense resistor  32 . The ramping current is filtered by an output capacitor  36  and supplies current to the load  38 . The output capacitor  36  is relatively large to smooth out ripple. 
         [0006]    The output voltage Vo is applied to a voltage divider  42 , and the divided voltage is applied to the negative input of a transconductance error amplifier  44 . Note this amplifier  44  can be either a current-output type transconductance amplifier or a voltage-output type amplifier. Capacitors may be connected across the resistors in the divider  42  to further compensate the feedback voltage. A reference voltage Vref is applied to the positive input of the amplifier  44 . The output current of the amplifier  44  corresponds to the difference between the actual output voltage Vo and the desired output voltage. The voltage (a control voltage Vc) across a capacitor  46  at the output of the amplifier  44  is adjusted up or down based on the positive or negative current output of the amplifier  44 . The control voltage Vc at the capacitor  46 , among other things, sets the duty cycle of the switch  26 , and the level of the control voltage Vc is that needed to equalize the inputs into the amplifier  44 . A resistor and capacitor may be connected in parallel with the capacitor  46  for controlling and optimizing the phase and loop stability, as is well known. 
         [0007]    The control voltage Vc is applied to a pulse width modulation (PWM) comparator  50 . The ramping voltage across the sense resistor  32 , when the switch  26  is on, is sensed by a differential amplifier  52 , having a certain gain, and, when the output of the amplifier  52  exceeds the control voltage Vc, the PWM comparator  50  is triggered to output a reset signal to the RS flip flop  20 . This turns the switch  26  off and turns the synchronous rectifier switch  28  on to discharge the inductor L 1 , causing a downward ramping current. In this way, the peak current through the inductor L 1  for each cycle is regulated to generate a desired output voltage Vo. The current through the sense resistor  32  includes a DC component (the lower frequency, average current) and an AC component (the higher frequency, ripple current). 
         [0008]      FIG. 1  also illustrates a conventional slope compensation circuit  59 , as is well known for current mode power converters. At high duty cycles (typically greater than 50%), the slope compensation circuit  59  turns off the switch  26  before the inductor current ramp crosses the control voltage Vc to reduce sub-harmonic oscillations that may occur in the current loop at the high duty cycles. The effect of the slope compensation circuit  59  is unrelated to the present invention. 
         [0009]    As will be described with respect to  FIG. 3 , switching noise (e.g., high frequency spikes and oscillations) by the turning on or off of the switch  26  is coupled to the current sense circuit and causes false triggering of the PWM comparator  50 , resulting in jitter and an increase of ripple on the output voltage Vo. 
         [0010]    The voltage drop and the power dissipation across the low value sense resistor  32  becomes more and more significant with higher currents and lower output voltages. It is desirable to use a small value sense resistor to reduce its power dissipation. Unfortunately, providing a very low value sense resistor  32  results in a low signal to noise ratio of the sensing signal, causing imprecise switching, in addition to the switching noise problem. 
         [0011]    Furthermore, it is desirable to even eliminate the sense resistor altogether to save power loss and improve the converter efficiency. 
         [0012]    Instead of detecting the inductor current through a sense resistor, the current through the inductor L 1  may be sensed by detecting the voltage drop across the switch  26  (e.g., a MOSFET). The on-resistance of such a MOSFETs may be a few mohms. However, such sensing still results in a low signal to noise ratio of the sensing signal and imprecise switching, in addition to the switching noise problem. 
         [0013]      FIG. 2  illustrates using the inherent DC winding resistance (DCR) of the inductor L 1  to detect the inductor current. An inductor winding may have a DC resistance on the order of a few mohms to less than 1 mohm. An RC network, comprising the series connection of a resistor R and capacitor C, connected across the inductor L 1  is selected to have substantially the same time constant as that of the inductor and DCR so that RC=L 1 /DCR. Accordingly, the ramping voltage across the capacitor C will track the ramping current through the inductor L 1 . The voltage across the capacitor C is then sensed by the differential amplifier  52 , and the remainder of the operation is the same as that described with respect to  FIG. 1 . The sensed voltage across the capacitor C includes a DC component (corresponding to the lower frequency, average current) and an AC component (corresponding to higher frequency, ripple current). In an application with very low inductor DCR value, the converter of  FIG. 2  suffers from the same switching noise problem and signal to noise ratio problem as described with respect to  FIG. 1 . Since the RC time constant must match the L 1 /DCR time constant for proper operation, the signal to noise ratio cannot be improved using the technique of  FIG. 2 . 
         [0014]      FIG. 3  illustrates the problem with switching noise. The clock pulse  62  (Clk in  FIGS. 1 and 2 ) turns on the switch  26  and turns off the switch  28 . The switching causes a high frequency oscillation due to the various parasitic capacitances and inductances in the system. When the sensed inductor current signal rises to cross the control voltage Vc, triggering the PWM comparator  50 , the switch  26  is turned off, creating switching noise. The resulting spike and oscillation can cause false triggering of the comparator  50 , resulting in a jittering of the comparator  50  output. This jitter is shown by the variability  63  in the on-time  64  of the switch  26 . This adversely affects the duty cycle control precision and the regulation of the output voltage Vo. The problem can become much worse in a multi-phase paralleled converter in which switching noises can be coupled among phases. 
         [0015]    What is needed is a current sensing technique for a switching power supply that reduces the jitter stemming from switching noise and also improves the signal to noise ratio of the current sense feedback loop with very low resistance value current sensing elements. 
       SUMMARY 
       [0016]    In a current mode controlled switching power supply, current through the inductor is sensed, by a current feedback loop, to determine when to turn off the switching transistor. A low resistance value current sensing element is preferred to minimize the power dissipation in the sensing element. The current feedback loop in the preferred embodiment of the invention both increases the signal to noise ratio of the ramping current sense signal and reduces the effect of the switching noise on the duty cycle control. The DC (lower frequency, average current) component of the sensed current and the AC (higher frequency, ripple current) component of the sensed current are measured using separate paths. The AC path has a higher signal to noise ratio than the DC path (for more precise detection of the ramping current level), and the DC path includes a low pass filter to filter out switching noise. The gain of the DC sense signal is adjusted upward so that the DC sense signal has the proper proportion to the AC sense signal to accurately reproduce the entire inductor current signal at the input of the PWM comparator. 
         [0017]    In one embodiment, for the AC sensing path, a first RC circuit connected across the inductor (L 1 ) has a time constant that is lower than L 1 /DCR so that the capacitor charges to a higher AC voltage compared to the prior art example of  FIG. 2 , where the time constant of the RC circuit was required to match L 1 /DCR. This enables more accurate detection of when the sensed current ramp crosses the control voltage Vc. For the DC path, a second RC circuit is connected across the inductor and has a time constant that is equal to or greater than L 1 /DCR (signal to noise ratio not improved). The DC sense signal is applied to a low pass filter to further eliminate the switching noise. The gain of the DC sense signal is controlled by a low voltage offset amplifier so that the DC sense signal is in the proper proportion to the AC sense signal so there is no distortion. The AC sense signal and DC sense signal are then summed. The summed signal is applied to the PWM comparator for comparison with the control voltage Vc. The summed signal has a high signal to noise ratio (due to the AC path) and reduced switching noise (due to the DC path). Thus the duty cycle is more precisely controlled. The amplifier may itself act as the low pass filter by using a capacitor in a negative feedback loop, or the filter may be a separate component. 
         [0018]    In another embodiment, the DC sense signal is sampled and held, further reducing the effect of switching noise. 
         [0019]    In another embodiment, the AC sense signal is detected and generated by measuring the voltage across the inductor, and the DC sense signal is detected as described in the other examples. The DC sense signal is gain-adjusted and summed with the AC sense signal. 
         [0020]    In another embodiment, the AC sense signal is detected by the first RC circuit as described in the other examples, and the DC sense signal is taken across the capacitor in the first RC circuit and then filtered to remove the switching noise. The DC sense signal is gain-adjusted and summed with the AC sensed signal. 
         [0021]    The DC or AC sense signal may also be detected across a separate sense resistor or across the power switch. 
         [0022]    The DC and/or AC sense signal may be processed digitally using analog-to-digital converters and a digital summing circuit or method. 
         [0023]    In the various examples, the switching noise is virtually eliminated in the DC path, due to the low pass filter or sample and hold circuit, prior to summing the AC and DC sense signals so as to reduce the effect of switching noise in the summed current sense signal. To further improve the performance, the signal to noise ratio of the AC path is also increased. 
         [0024]    Various other embodiments are described. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0025]      FIG. 1  illustrates a prior art DC/DC converter using a sense resistor to detect inductor current. 
           [0026]      FIG. 2  illustrates another prior art DC/DC converter using the DCR of the inductor winding to detect inductor current. 
           [0027]      FIG. 3  illustrates the effect of switching noise on switch on-time jitter. 
           [0028]      FIG. 4  illustrates a first embodiment of the invention using different RC circuits for the AC sense path and the DC sense path. 
           [0029]      FIG. 5  illustrates a second embodiment of the invention using a sample and hold circuit in the DC sense path. 
           [0030]      FIG. 6  illustrates a third embodiment of the invention where the AC current signal is sensed and generated by detecting the voltage across the inductor. 
           [0031]      FIG. 7  illustrates a fourth embodiment of the invention where the AC and DC paths use the same RC circuit, and the DC sense signal is filtered by a low pass filter to filter out the switching noise and the overly amplified AC signal. 
           [0032]      FIG. 8  illustrates a fifth embodiment of the invention where the DC current is sensed across a sense resistor. 
           [0033]      FIG. 9  illustrates a sixth embodiment of the invention where the DC current is sensed across a synchronous rectifier and held during the power switch on-time using a sample and hold circuit. 
           [0034]      FIG. 10  illustrates a seventh embodiment of the invention where the AC and DC path signal processing is performed digitally. 
       
    
    
       [0035]    Elements that are the same or equivalent are labeled with the same numeral. 
       DETAILED DESCRIPTION 
       [0036]    In the various embodiments of the invention shown in  FIGS. 4-10 , only the aspects of the converter that are different from the converter of  FIG. 1  are shown for simplicity. Accordingly, the output of the PWM comparator  50  in  FIGS. 4-10  is coupled to the switching circuitry as shown in  FIG. 1 , the output terminal of the inductor L 1  is connected to the output circuit of  FIG. 1 , and the voltage feedback path for generating the control voltage Vc is that shown in  FIG. 1 . Other suitable circuitry may be used in conjunction with the present invention instead of the circuitry shown in  FIG. 1 . 
         [0037]      FIG. 4  illustrates a first embodiment of the invention using different RC circuits for the AC sense path and the DC sense path. A resistor R 1  and capacitor C 1  are connected in series across the inductor L 1 . The resistance DCR represents the inductor winding DC resistance. In contrast to the prior art  FIG. 2 , where the RC time constant should match the L 1 /DCR time constant to obtain accurate current sensing, the R 1 *C 1  time constant is significantly less than L 1 /DCR to generate an enlarged AC ripple signal and therefore increase the signal to noise ratio of the AC voltage across capacitor C 1 . All the AC voltages are sensed across the capacitor C 1  terminals, although only one AC lead is shown in the examples for simplicity. The R 1 *C 1  time constant may be any time constant below L 1 /DCR for proper operation, since the gain of the DC path is adjusted to avoid distortion. 
         [0038]    Either the R 1  value or the C 1  value or both may be reduced from that of  FIG. 2  to lower the time constant. By lowering the time constant of R 1 *C 1 , the AC ripple voltage magnitude across capacitor C 1  can be greatly increased compared to that in  FIG. 2 , as the current through the inductor L 1  ramps up. This increased voltage signal increases the signal to noise ratio of the AC sense signal, making triggering of the PWM comparator  50  more accurately timed. 
         [0039]    Since any switching noise contains frequencies much greater than the switching frequency, much of the switching noise will be filtered out by the capacitor C 1 , so the reduction of the time constant of R 1 *C 1  does not adversely impact the effect of switching noise in the AC path. 
         [0040]    A second current sensing path is formed by the series connection of resistor R 2  and capacitor C 2  across the inductor L 1 . The time constant of R 2 *C 2  is equal to or greater than L 1 /DCR, but preferably greater than so as to further reduce the effects of ripple and noise in the DC path. The voltage across the capacitor C 2  is applied to the differential inputs of a low offset voltage (Vos) differential amplifier  68  having a gain of K. The output of the amplifier  68  is applied to a low pass filter  70 . The filter  70  filters out virtually all of the high frequency switching noise. The DC sense voltage is K*Vsen(dc). 
         [0041]    The low pass filter  70  may be a capacitor connected to ground or may be a capacitor in a negative feedback path in the differential amplifier  68 . Therefore, the low pass filter  70  is drawn in dashed outline. 
         [0042]    In all the drawings showing a separate low pass filter and amplifier in the DC path, the filter and amplifier relative positions may be reversed. A differential filter would then be used. 
         [0043]    The AC sense signal (Vsen(ac)) and the DC sense signal (K*Vsen(dc)) are summed by a conventional summer  72  to generate a composite current sense signal (k1*Vsense), where k1 is the total signal gain of the combined current sense signal, and Vsense is the actual voltage across the DCR. The value of (k1*Vsense) will be proportional to i L *DCR, where i L  is the current through the inductor L 1 . The gain of the amplifier  68  (greater than 1) is set such that the DC sense signal has the proper proportion to the AC sense signal to accurately convey the current through the inductor L 1 . A decreased time constant R 1 *C 1  requires an increased gain of the amplifier  68  due to the increased AC ripple voltage across C 1 . The proper gain may be determined by simulation or frequency domain analysis. 
         [0044]    Accordingly, the composite current sense signal has a higher signal to noise ratio compared to that of  FIG. 2  and has less switching noise and jitter. 
         [0045]    The composite current sense signal may also be used for current limiting, current sharing, and other uses. This technique may also be used in a phased converter, where each phase generates a portion of the output current. 
         [0046]    In the various embodiments, although the term “DC” is used to identify one of the paths, the DC signal may vary at a relatively low frequency, representing an average current, as the load current varies. The terms DC and AC are intended to distinguish between the two paths and not intended to limit them. 
         [0047]      FIG. 5  illustrates a second embodiment of the invention, similar to  FIG. 4  but using a sample and hold circuit  76  in the DC sense path. The sample and hold circuit  76  further reduces noise and ripple by sampling the voltage across the capacitor C 2  at a time when the voltage is a midpoint of the voltage ramp, representing an average current. The sampling clock pulse is triggered when the sample sensor  78  detects when the voltage across the capacitor C 2  is midway between its two peaks. This sensing may be implemented using known techniques. The sampled signal is held until the start of the next switching cycle. Accordingly, switching noise is eliminated from the DC path. In one embodiment, the low pass filter  70  is not used when the DC signal is sampled. 
         [0048]      FIG. 6  illustrates a third embodiment of the invention where the AC current is sensed by directly detecting the voltage (Vsw−Vo) across the inductor. The di/dt through the inductor L 1  is (Vsw−Vo)/L 1 . When the switch  26  ( FIG. 1 ) is on, Vsw will be approximately the input voltage Vin. The Vsw and Vo voltages are applied to the inputs of a transconductance amplifier  80 . The current output by the amplifier  80  charges a capacitor  82  to generate a varying sense voltage Vsen(ac). The capacitor  82  value can be reduced to enlarge the AC ripple signal to increase the signal to noise ratio. The R 2 C 2  circuit may be identical to that shown in  FIG. 4 , except the gain of the amplifier  68  may be different to create an accurate composite signal. Accordingly, the signal to noise ratio in the AC path is increased, and the switching noise is lowered in the DC path, to create a more precise converter. 
         [0049]      FIG. 7  illustrates a fourth embodiment of the invention where the AC and DC paths use the same R 1 C 1  circuit, and the DC sense signal is filtered by a low pass filter  70  to filter out the switching noise and AC ripple. The R 1 C 1  circuit is similar to that of  FIG. 4 , where the time constant is less than that of L 1 /DCR to obtain a higher signal to noise ratio. The AC sense signal is taken across the capacitor C 1  as in  FIG. 4 . The DC sense signal is obtained by detecting the voltage across the capacitor C 1 , then filtering the signal by the low pass filter  70  to remove switching noise and AC ripple, then amplifying the signal by the amplifier  68  to cause the DC sense signal to have the proper proportion to the AC sense signal for no distortion. As previously mentioned, the amplifier  68  may also perform the filtering function. An advantage of the circuit of  FIG. 7  is that an IC package that houses the control circuit uses only two pins to access the external C 1  terminals for current sensing, just like the two terminals needed in the prior art controllers of  FIGS. 1 and 2 . Accordingly, the same packages can be used. Further, only one RC network is needed externally for current sensing. 
         [0050]      FIG. 8  illustrates a fifth embodiment of the invention where the DC current is sensed across a low value sense resistor Rsense, which is typically formed to a tighter tolerance than the inductor DCR. The AC sense path is the same as in  FIG. 4  except R 1 C 1  is connected across the inductor L 1  and Rsense. The voltage detected across Rsense is applied to the amplifier  68  and then filtered to remove the switch noise and ripple. As in previous embodiments, the gain K of the amplifier  68  is set to cause the DC sense signal to have the correct proportion to the AC sense signal to obtain an accurate composite current sense signal. 
         [0051]      FIG. 9  illustrates a sixth embodiment of the invention where the DC current is sensed across a switch.  FIG. 9  is similar to  FIG. 8  in that the DC current is sensed across a resistance in series with the inductor L 1 . In this case, it is the on-resistance of the synchronous rectifier MOSFET  86 . A midpoint of the downward sloping current ramp, when the synchronous rectifier MOSFET  86  is on, will be the same as the midpoint of the upward sloping current ramp when the power MOSFET  88  is on. Therefore, a sample and hold circuit  76  is controlled by the sample sensor  78  to sample the voltage across the MOSFET  86  at the midway point and hold the voltage during the time the power MOSFET  88  is on. The sampled voltage, after being amplified and filtered, is then summed with the AC sense signal during the time the power MOSFET  88  is on to create the composite current sense signal. The switching noise and ripple are effectively removed by the sample and hold circuit  76 . In the embodiment, using a sample and hold circuit, the low pass filter  70  is optional. Similarly, the DC current signal can also be sensed across the top side power switch  88 , with a sample and hold circuit sensing the current at midpoint of the upward sloping inductor current ramp. 
         [0052]      FIG. 10  illustrates a seventh embodiment of the invention where the AC and DC path signal processing is performed digitally. Instead of pure analog sensing, amplifying, and summing in all the embodiments, the voltages detected are converted to digital signals by analog-to-digital converters (ADCs)  90  and  92  and then processed digitally. In one embodiment, the summing is performed digitally, and the comparison with the control voltage Vc (converted to a digital signal) is performed digitally. In such a case, the PWM comparator  50  is implemented as a digital comparator. The particular implementation of  FIG. 10  is just an example of how any of the embodiments can be converted to perform various processes in the digital domain. 
         [0053]    The various embodiments described herein may be combined in any way such that there are separate AC and DC sense paths, where the DC path has switching noise and ripple removed and/or the AC path has an increased signal to noise ratio. Additionally, although an amplifier with a gain greater than 1 has been shown in the DC path to adjust the magnitude of the DC sense signal to have the proper proportion to the AC sense signal, the amplifier may instead be inserted into the AC path, with a gain less than one. 
         [0054]    While particular embodiments of the present invention have been shown and described, it will be obvious to those skilled in the art that changes and modifications may be made without departing from this invention in its broader aspects and, therefore, the appended claims are to encompass within their scope all such changes and modifications that are within the true spirit and scope of this invention.