Abstract:
A heterodyne optical signal detector and method performed thereby, the signal detector including an optical signal spectrum shaper operable to modify the shape of the frequency spectrum of a received optical signal, a laser local oscillator (LO), and polarization beam splitters (PBSs) to divide the signal and the LO into orthogonal components, waveguides in which intermediate frequency (IF) signals are formed, balanced photodetectors (BPDs) arranged to receive the IF signals and operable to convert the IF signals into electric signals, and analog to digital converters (ADCs) that digitize the electric signals. In embodiments, the ADCs have a predetermined bandwidth, the received signal has a spectrum which, if not shaped, would produce IF signals with a bandwidth greater than that of the ADCs, the spectrum shaper modifies the received signal spectrum to produce IF signals that have a bandwidth substantially equal to half the bandwidth of the ADCs, the laser frequency is adjusted produce IF signals in the waveguides having a frequency spectrum centered at the midpoint of the ADCs&#39; bandwidth.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This Application claims the benefit of U.S. Provisional Application 61/709,496 filed on Oct. 4, 2012. 
     
    
     BACKGROUND 
       [0002]    With the development of large-bandwidth and high-speed electronic analog-to-digital converters (ADCs) and photo detectors (PDs), coherent detection with digital signal processing (DSP) has been attracting a great deal of interest in research community. It is well known that coherent detection can include either homodyne detection or heterodyne detection. However, unlike homodyne detection, heterodyne detection can simultaneously down-convert in-phase (I) and quadrature (Q) components to an intermediate frequency (IF), thereby reducing the number of balanced PDs and ADCs of a coherent receiver by half. Furthermore, with heterodyne detection there is no need to consider the delays between the I and Q components in a polarization-division-multiplexed (PDM) signal. In addition, with heterodyne detection a conventional dual-hybrid structure is also unnecessary. Accordingly, heterodyne detection is much more hardware-efficient than homodyne detection. 
         [0003]    However, the ADC bandwidth needed for heterodyne detection is twice that needed for homodyne detection. It is well known that in heterodyne detection there exists a frequency offset, i.e., the frequency difference between the local oscillator (LO) source and a received optical signal. Thus, in the case where the ADC bandwidth is limited and the signal spectrum is wide (e.g., larger than the ADC bandwidth), the prior art does not optimize the frequency offset for heterodyne detection, resulting in undesirable signal spectrum overlap and/or cutoff. 
       SUMMARY 
       [0004]    A heterodyne optical signal detector and method performed thereby, the signal detector including an optical signal spectrum shaper operable to modify the shape of the frequency spectrum of a received optical signal, a laser local oscillator (LO), and polarization beam splitters (PBSs) to divide the signal and the LO into orthogonal components, waveguides in which intermediate frequency (IF) signals are formed, balanced photodetectors (BPDs) arranged to receive the IF signals and operable to convert the IF signals into electric signals, and analog to digital converters (ADCs) that digitize the electric signals. In embodiments, the ADCs have a predetermined bandwidth, the received signal has a spectrum which, if not shaped, would produce IF signals with a bandwidth greater than that of the ADCs, the spectrum shaper modifies the received signal spectrum to produce IF signals that have a bandwidth substantially equal to half the bandwidth of the ADCs, the laser frequency is adjusted produce IF signals in the waveguides having a frequency spectrum centered at the midpoint of the ADCs&#39; bandwidth. 
         [0005]    It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are intended to provide further explanation of the invention as claimed. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0006]    The accompanying drawings are included to provide a further understanding of the invention, and are incorporated in and constitute a part of this specification. The drawings illustrate disclosed embodiments and/or aspects and, together with the description, serve to explain the principles of the invention, the scope of which is determined by the claims. 
           [0007]    In the drawings: 
           [0008]      FIG. 1  inset A is a block diagram of a simplified heterodyne coherent receiver, and inset B illustrates IF down conversion in digital frequency domain; 
           [0009]      FIG. 2  shows optimal frequency offset and spectrum shaping for heterodyne coherent detection in accordance with the disclosure; 
           [0010]      FIG. 3  illustrates the evolution of 9-QAM signal generation by digital post filtering; 
           [0011]      FIG. 4  is a block diagram of an exemplary setup for the generation and heterodyne detection of a 220-Gb/s single-channel PDM-QPSK signal on a 50-GHz grid with post filter and 1-bit MLSE, in accordance with the disclosure; 
           [0012]      FIG. 5  shows optical spectra before and after spectral shaping by a programmable wavelength selective switch, in accordance with the disclosure; and 
           [0013]      FIG. 6  inset A is a graph of back-to-back (BTB) bit error rate (BER) versus optical signal to noise ratio with and without post filter and 1-bit MLSE in accordance with the disclosure, and inset B is a graph of BTB BER versus frequency offset when the OSNR is set at 32 dB. 
       
    
    
     DETAILED DESCRIPTION 
       [0014]    It is to be understood that the figures and descriptions provided herein may have been simplified to illustrate elements that are relevant for a clear understanding of the present invention, while eliminating, for the purpose of clarity, other elements found in typical optical signal detection systems and methods. Those of ordinary skill in the art may recognize that other elements and/or steps may be desirable and/or necessary to implement the devices, systems, and methods described herein. However, because such elements and steps are well known in the art, and because they do not facilitate a better understanding of the present invention, a discussion of such elements and steps may not be provided herein. The present disclosure is deemed to inherently include all such elements, variations, and modifications to the disclosed elements and methods that would be known to those of ordinary skill in the pertinent art. 
       Principle of Simplified Heterodyne Detection and Digital Post Filtering 
     Simplified Heterodyne Detection 
       [0015]      FIG. 1A  is a block diagram illustrating an exemplary heterodyne coherent receiver, and  FIG. 1B  illustrates intermediate frequency (IF) down conversion in the digital frequency domain. In the figure, inset A shows a simplified heterodyne coherent receiver. Inset B illustrates IF down conversion in the digital frequency domain. In the figure, the following labels are used. LO: local oscillator, PBS: polarization beam splitter, OC: optical coupler, BPD: balanced photo detector, ADC: analog-to-digital converter. 
         [0016]    The heterodyne coherent receiver includes two polarization beam splitters (PBSs) for polarization-diversity splitting between the received optical polarization-division-multiplexing (PDM) signal and the local oscillator (LO) source, two optical couplers (OCs), two balanced photo detectors (PDs) and two analog-to-digital converters (ADCs). Only two balanced PDs and two ADCs are needed for the heterodyne detection. The polarization-diversity hybrid is also simplified in comparison to a conventional hybrid for homodyne detection. As used herein, the term “intermediate frequency” (IF) denotes the frequency offset from a received signal frequency. In-phase (I) and quadrature (Q) signal components are received simultaneously, centered at the IF, as shown in  FIG. 1B  wherein f IF  denotes the IF, and B W  the bandwidth of the I or Q component. In order to separate the I and Q components without crosstalk, f IF ≧B W  should be satisfied. The IF can be down-converted to the baseband frequency in the digital frequency domain. After polarization-diversity splitting, the received optical signal of the X-polarization state can be expressed as 
         [0000]        E   S ( t )=√{square root over ( P   S )}exp[ j 2 πf   S   t+φ   S ( t )].  (1)
 
         [0000]    where P S , f S  and φ S  are the power, carrier frequency, and phase, respectively, of the received optical signal of X-polarization state. Similarly, the LO source of the X-polarization state can be expressed as 
         [0000]        E   LO ( t )=√{square root over ( P   LO )}exp[ j 2 πf   LO   t+φ   LO ( t )].  (2)
 
         [0000]    where P LO , f LO  and φ LO  are the power, carrier frequency, and phase of the LO source of X-polarization state, respectively. 
         [0017]    f the bandwidth of the ADCs and PDs is large enough, after balanced photodetection the generated electrical signal of X-polarization state consists of both the baseband and IF components carrying the entire I and Q components. This is expressed as 
         [0000]        I   BPD ( t )= P   S   +P   LO +2 R √{square root over ( P   S   P   LO )}exp[ j 2 πf   IF   t+φ   IF ( t )].
 
         [0000]        f   IF   =f   S   −f   LO ; 
         [0000]      φ IF ( t )=φ S ( t )−φ LO ( t ).  (3)
 
         [0000]    where R is the PD responsivity, and f IF  and φ IF  denote the frequency and phase of the IF component, respectively. The output current of balanced PD for Y-polarization state is similar to Eq. 3. By multiplying a proper transfer function, the extracted IF component can be expressed as 
         [0000]        E   IF ( t )=2 R √{square root over ( P   S   P   LO )}·[ I ( t )·cos(2 πf   IF   t+φ   IF ( t ))+ Q ( t )·sin(2 πf   IF   t+φ   IF ( t ))].  (4)
 
         [0000]    where I(t) and Q(t) denote the I and Q components of the received optical PDM signal, respectively. By re-choosing the zero-frequency point of Eq. (4), the IF component can be down-converted to the baseband by simple frequency shifting. Compared to the external IF down conversion based on frequency beating with electrical mixer and radio-frequency (RF) signal, basic operation in the digital frequency domain is much more hardware-efficient. 
         [0018]    Assume ζ is the noise density of a signal. The signal-to-noise ratio (SNR) for heterodyne detection is 
         [0000]    
       
         
           
             
               
                 
                   
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         [0019]    In contrast, for homodyne detection satisfying f IF =0, a conventional 2×4 optical 90° hybrid is needed. In that case, the I or Q component of the signal can be expressed as 
         [0000]        I   BPD     —     i/q =2 R √{square root over ( P   S   P   LO  )}cos [φ S ( t )−φ LO ( t )].  (6)
 
         [0020]    As a result, the SNR for homodyne detection is 
         [0000]    
       
         
           
             
               
                 
                   
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         [0021]    Thus, homodyne detection has a 3-dB SNR improvement compared to heterodyne detection. But homodyne detection cannot use the simplified optical 180° hybrid described previously with regard to heterodyne detection, and cannot use only two balanced PDs and two ADCs without destroying I and Q information. 
         [0000]    Optimal Frequency Offset and Spectrum Shaping with Digital Post Filtering 
         [0022]      FIG. 2  illustrates the effects of frequency offset and spectrum shaping on heterodyne coherent detection. Due to the bandwidth limit of an ADC in practice, the frequency offset between the LO signal and the received signal should not be too large. As shown in  FIG. 2A , the signal spectrum beyond the ADC bandwidth (B ACD ) may be cut off when a large frequency offset (f IF ) is used. To prevent such cutoff, the signal spectrum may be shaped to fall within the ADC bandwidth. As shown in  FIG. 2B , both spectrum overlap and cut-off may occur when the signal is not spectrum shaped and is larger than the ADC bandwidth. Accordingly, spectrum shaping can be used in conjunction with adjusting the frequency offset to achieve an optimal high speed heterodyne coherent detection.  FIG. 2C  illustrates an optimized frequency offset and spectrum shaped signal. As shown, when the frequency offset and spectrum shape are optimized, the received signal uses the entire available ADC bandwidth without any cut off of the signal spectrum beyond the B ADC , and without any spectrum overlap. As shown, the spectrum shape is optimized when the shaped signal bandwidth is half the ADC bandwidth, and the frequency offset is optimized when the shaped signal spectrum utilizes the entire ADC bandwidth with no spectrum cut off or overlap. 
         [0023]    Duo-binary signaling or correlative coding is a specific class of partial response signaling that has only 1-bit memory length, and that introduces a controlled amount of inter-symbol interference (ISI) into the signal, rather than trying to eliminate ISI as is common. The introduced ISI can be compensated for by using digital signal processing in the digital domain at the receiver. The ideal symbol-rate packing of 2 symbols per Hertz can then be achieved without encountering the requirements for unrealizable optical filters based on the Nyquist theorem. Multi-symbol optimal decision schemes, such as maximum-a-posteriori probability (MAP) estimation, maximum likelihood sequence estimation (MLSE), and the like, can take advantage of symbol correlation that exists in received partial response signals. The challenge is that the number of states and transitions grows exponentially with increasing memory length. For instance, an adopted MLSE length of 10 results in 410 states and 411 transitions in lane-dependent PDM-QPSK signals. Consequently, computational complexity significantly increases in practical implementations. 
         [0024]    Moreover, in a bandwidth-limiting optical coherent system, noise in high frequency components of the signal spectrum and inter-channel crosstalk may both be made worse by using a conventional linear equalization algorithm, such as the conventional constant modulus algorithm (CMA), for example. However, in embodiments, a linear digital delay-and-add finite impulse response (FIR) post filter provides a simple way to achieve partial response that can effectively mitigate the enhanced inter-channel crosstalk and intra-channel noise introduced by an adaptive equalizer. At the coherent receiver, the post filter is added after carrier phase estimation (CPE) in the conventional DSP process. From the constellation point of view, the effect of the post filter transforms the 4-point QPSK signal into 9-point quadrature duo-binary one. The evolution of this transformation is illustrated in  FIG. 3 . As a result of the delay-and-add effect, the 2-ary amplitude shift keying (2-ASK) I and Q components disappear and are then independently converted into two 3-ASK symbol series. The mechanism for the generation of ‘9-QAM’ signals can be considered as the superposition of the two 3-ASK vectors on a complex plane. In  FIG. 3 , the size of constellation points represents the relative number of points generated after the post filter. 
         [0025]    The adoption of the post filter also makes possible the use of MLSE with just a 1-bit memory length, which can effect further error correction that may be induced by ISI. 
         [0000]    Optimal Frequency Offset in a 55-Gbaud PDM-QPSK Single Channel on a 50-GHz Grid with Simplified Heterodyne Detection 
         [0026]      FIG. 4  shows an exemplary setup for the generation, 405, and heterodyne detection, 410, of a 220-Gb/s single-channel PDM-QPSK signal on a 50-GHz grid with post filter and 1-bit MLSE. In  FIG. 4 , inset A shows the electrical spectra for the Y-polarization component centered on 30, 28, and 25 GHz obtained after balanced detection. Inset B shows the detailed DSP after analog-to-digital conversion. Inset C shows the received constellations corresponding to 25-GHz frequency offset after CPE and further post filtering for the X-polarization state, while inset D shows the Y-polarization state. In the figure, the following labels are used. ECL: external cavity laser, PPG: pulse pattern generator, I/Q MOD: I/Q modulator, EA: electrical amplifier, OC: optical coupler, DL: delay line, ATT: optical attenuator, PBC: polarization beam combiner, EDFA: Erbium-doped fiber amplifier, WSS: wavelength selective switch, PBS: polarization beam splitter, LO: local oscillator, BPD: balanced photo detector, OSC: oscilloscope. 
         [0027]    At transmitter  415 , a continuous-wavelength (CW) lightwave at 1549.34 nm from external cavity laser (ECL)  420 , with linewidth less than 100 kHz and maximum output power of 14.5 dBm, is modulated by an I/Q modulator (I/Q MOD). The I/Q modulator is driven by a 55-Gbaud electrical binary signal, which, with a pseudo-random binary sequence (PRBS) length of 215-1, is generated from an electrical dual-channel pulse pattern generator (PPG). For optical QPSK modulation, two parallel Mach-Zehnder modulators (MZMs) in the I/Q modulator are both biased at the null point and driven at the full swing to achieve zero-chirp 0-and Tr-phase modulation. The phase difference between the upper and lower branches of the I/Q modulator is controlled at π/2. Subsequent polarization multiplexing is effected by polarization multiplexer 425, comprising a polarization-maintaining optical coupler (00) to halve the signal into two branches, an optical delay line (DL) to provide a 150-symbol delay, an optical attenuator (ATT) to balance the power of the two branches, and a polarization beam combiner (PBC) to recombine the signal. The signal is then power-amplified by an Erbium-doped fiber amplifier (EDFA) for transmission. Thereafter, the 220-Gb/s PDM-QPSK single channel is spectrally shaped by a programmable wavelength selective switch (WSS) on a 50-GHz grid. 
         [0028]    At the receiver, another laser (ECL2) with linewidth less than 100 kHz is used as the LO source, which has a large frequency offset relative to the received optical signal. Two polarization beam splitters (PBSs) and two OCs are used to realize polarization diversity of the received signal with the LO source in optical domain before balanced PDs each with 50-GHz bandwidth. Analog-to-digital conversion is realized in a real-time digital storage oscilloscope (OSC) with 120-GSa/s sampling rate and 45-GHz electrical bandwidth. Two ADC channels provide for offline DSP. 
         [0029]    In the receiver, the received signals are down-converted to the baseband by frequency shifting. The digital IF down conversion is described in detail hereinafter. A T/2-spaced time-domain FIR filter is then used for CD compensation, where the filter coefficients are calculated from the known fiber CD transfer function using the frequency-domain truncation method. I and Q components are then separated by multiplying synchronous cosine and sine functions, which are generated from a digital LO for down conversion. Then, two complex-valued, 13-tap, T/2-spaced adaptive FIR filters, based on classic CMA, are used to retrieve the modulus of the PDM-QPSK signal and realize polarization de-multiplexing. The subsequent step is carrier recovery, which includes residual frequency-offset estimation and CPE. The former is based on a fast Fourier transform (FFT) method, while the latter is based on a fourth-power Viterbi-Viterbi algorithm. A post filter is then adopted to convert the binary signal to a duo-binary one. The final symbol decision is based on a 1-bit MSLE. Finally, differential decoding is used to eliminate π/2 phase ambiguity before bit-error rate (BER) counting. In this exemplary embodiment, the BER is counted over 10×106 bits (10 data sets, and each set contains 106 bits). 
         [0030]      FIG. 5  shows the optical spectra (0.02-nm resolution) for the PDM-QPSK single channel at 1549.34 nm before A and after B the 50-GHz WSS. It can be seen that the optical spectrum becomes much narrower after WSS. 
         [0031]    Referring now to  FIG. 6 , for the exemplary 220-Gb/s PDM-QPSK single channel at 1549.34 nm, inset A shows the measured back-to-back (BTB) BER versus the optical SNR (OSNR) with and without the technique of post filter and 1-bit MLSE, while inset B shows the measured BTB BER versus the frequency offset when the OSNR is set at 32 dB. It can be seen from inset A that, after adopting post filter and 1-bit MLSE, the BER performance is much better and the BER is less than the pre-forward-error-correction (pre-FEC) limit of 3.8×10−3 when the OSNR is over 25.6 dB. From inset B, it can be seen that the optimum BER performance is attained when the frequency offset is 22˜23 GHz, i.e., half of the ADC bandwidth. 
         [0032]    The herein disclosed apparatus, systems, and methods can be used to detect optical signals that were generated using any type of phase-shift keying (PSK) or quadrature amplitude modulation (QAM) modulation scheme, such as 4PSK, 8PSK, 16PSK, 4QAM, 8QAM, 16QAM, 64QAM, 256QAM, 1024QAM, 4096QAM, or higher order schemes, Offset Quadrature PSK (OQPSK), Differential PSK (DPSK), or any other variant of PSK or QAM. 
         [0033]    Although the invention has been described and illustrated in exemplary forms with a certain degree of particularity, it is noted that the description and illustrations have been made by way of example only. Numerous changes in the details of construction and combination and arrangement of parts and steps may be made. Accordingly, such changes are intended to be included in the invention, the scope of which is defined by the claims.