Abstract:
The invention relates to an auxiliary switching circuit ( 10 ) for a chopping converter comprising a first inductive element (L 0 ) for serial energy storage with a free-wheel diode (DL) and a switch (K), in addition to a second inductive element (L) for di/dt control when the switch is closed, the auxialiary switching circuit comprising a magnetic circuit ( 11 ) whereby a main winding thereof is formed at least partially by the first inductive element (L 0 ), also comprising means (L 1 , D 1 , L 2 , D 2 ) for discharging the second inductive element when the switch is opened or closed, and means (L 2 , D 2 ) for transferring the energy corresponding to the closure vis a vis said main winding.

Description:
BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to the field of power converters of switched-mode type. Such converters use an inductive element, associated with a power switch and with a free wheel diode, to perform a power conversion and a correction of the power factor, generally based on a D.C. input voltage. Voltage step-down converters (BUCK), voltage step-up converters (BOOST), and buck-boost converters are known. 
   The present invention more specifically relates to a circuit for helping the switching of the power switch of a switched-mode converter. 
   2. Discussion of the Related Art 
     FIG. 1  shows the simplified diagram of a conventional step-up converter  1 . Such a converter includes an inductance L 0  in series with a free wheel diode DL between two positive input and output terminals  2  and  3  of the converter, the cathode of diode DL being connected to terminal  3 . A power switch K connects the midpoint  4  of this series connection to a terminal  5  of application of a negative or reference voltage (generally, the ground) common to the converter input and output. A D.C. supply voltage source  6  provides a voltage V E  across terminals  2  and  5 . On the output side, a storage capacitor C 0  generally connects terminals  3  and  5  and provides a voltage V S  to a load Q. Load Q has been shown in  FIG. 1  by dotted lines integrating capacitor C 0 , which may or not belong to the load. Switch K is controlled by a circuit  7  (CTRL), for example, in pulse-width modulation (PWM). 
   The operation of a step-up converter will now be described. When switch K is on, power is stored in inductance L 0  and load Q is supplied by the power stored in capacitor C 0 . When switch K is off, inductance L 0  gives back the stored power to capacitor C 0  via free wheel diode DL. 
     FIG. 2  shows the simplified electric diagram of a step-down converter  1 ′. It shows the same components as in  FIG. 1 . However, here, switch K is connected in series with inductance L 0  between positive input and output terminals  2  and  3 . Free wheel diode DL grounds the junction point  4 ′ of switch K and inductance L 0 , its cathode being connected to point  4 ′. Switch K may also be provided between the negative terminal of source  6  and the anode of diode DL. 
   The operating principle is the same. Power is stored in inductance L 0  during the on periods of switch K. During periods when switch K is off, this power is given back to capacitor C 0 , free wheel diode DL being used to loop back the circuit. 
   A problem which arises with switched-mode converters, also called hard-switching converters, in which the current and the voltage cross each other upon each switching, is linked to the switch turning-on. 
   Indeed, upon each turning-on of switch K, free wheel diode DL must block. Now, at the blocking of a diode, especially of a PN junction diode, a recovered charge phenomenon occurs. 
   This phenomenon is illustrated by  FIGS. 3A to 3C , which show, in relation with the circuit of  FIG. 1 , an example of the shape of current I DL  in the free wheel diode, of output voltage V S  and of current I T  in switch K. 
   Switch K is initially assumed to be off. Accordingly, a current I Lf  flows through diode DL. This current corresponds to the power given back by inductance L 0 . The output voltage is at a level V 0 . As for switch K, the current I T  flowing therethrough is null. 
   It is assumed that at a time t 1 , control circuit  7  turns switch K on. During the switching, current I L  in the inductance, which corresponds to the sum of currents I DL  and I T  is a constant. Accordingly, the current which, during the switching, increases in the switch, translates as a decrease with an inverse slope of the current in diode DL. 
   At a time t 2 , the current in diode DL becomes zero and the current in the switch reaches level I Lf . At this time starts the recovered charge phenomenon of diode DL. This known phenomenon translates as an inversion of the current through the diode to reach a level I RM  corresponding to the maximum recovery current of the diode. Current I RM  is reached at a time t 3  from which the current through the diode tends towards zero again, reaching it at a time t 4 . Since the current in inductance L 0  is, during the switching, substantially constant, the negative current peak on the diode side translates as an overcurrent in switch K, the maximum value of which corresponds to current I Lf  plus value I RM . On the side of voltage V S , the voltage decrease in practice intervenes from time t 3 , that is, from the inversion of the current slope in diode DL. In other words, the voltage across the diode is zero between times t 2  and t 3  corresponding to the first recovery phase ta. It can be considered that the diode then transiently conducts in reverse. Between times t 3  and t 4  (second recovery phase tb), voltage V S  decreases from V 0  to a zero voltage. The voltage provided to capacitor C 0  is here considered. Indeed, the presence of the capacitor in practice results in output voltage V S  remaining approximately stable. 
   The slope between times t 1  and t 3  of the current decrease in diode DL depends on the turn-on speed of the switch and thus on its di/dt at the turning-on. The higher this di/dt, which favors an abrupt switching, the higher amplitude I RM  is for a PN-junction diode. However, the smaller di/dt, the longer the recovery time at the blocking (trr=t 4 −t 2 ). 
   The losses in a diode according to the di/dt value have a parabolic shape. There is an optimal point where the surface area of the current shape between times t 2  and t 4  is minimum, which results in minimum losses of recovered charges in the diode. 
   For switch K, the recovered charge phenomenon of the diode is particularly disturbing. Indeed, for a step-up converter, the switch then sees across its terminals, between times t 2  and t 3 , output voltage V S . In the case of a step-down converter, the voltage seen by the switch across its terminals corresponds to the voltage of generator  6 . In all cases, it is the highest voltage between voltages V E  and V S . 
   High losses can then be observed in switch K. In  FIGS. 3A to 3C , the loss periods have been symbolized by hatching on the various timing diagrams. 
   In practice, the losses in switch K (generally, a power transistor) at its turning-on (times t 1  to t 4 ) form most of the switching losses of the converter. In particular, the losses due to the actual blocking of the diode and the turn-off losses of the switch are negligible with respect to the losses generated at its turning-on. 
   A first solution to reduce this disadvantage consists of using diodes with no recovered charges, for example, Schottky or SIC-type diodes. 
   A first disadvantage of this solution is that diodes with no recovered charges are often limited to a breakdown voltage of some hundred volts. This solution is thus not applicable to converters operating under voltages of several hundreds of volts, which is in practice current in power electronics. Several diodes in series must then be provided to increase the breakdown voltage. 
   Another disadvantage of this solution is that, even if it decreases losses linked to recovered charges (times t 2  to t 4 ), the most significant losses linked to the sole switch turning-on are not avoided. Referring to the example of  FIGS. 3A to 3C , the use of a diode with no recovered charges results in an zero voltage V S  from time t 2 . There thus remain the losses linked to the surface areas located between times t 1  and t 2 . 
   Another disadvantage of diodes with no recovered charges is that they are particularly expensive as compared to PN diodes. Presently, the cost ratio is greater than 20. 
   A second solution to attempt solving recovered charge problems is to provide a circuit for helping the switching of the power switch of the converter. 
     FIG. 4  shows a conventional example of such an aid circuit, applied to a step-up converter such as shown in  FIG. 1 .  FIG. 4  shows all elements of  FIG. 1 , to which is added a circuit  8  for helping the switching of switch K. This circuit is formed of an inductance L, associated in parallel with a resistor R and a diode D, between point  4  and switch K. The function of inductance L is to control the switch di/dt. By decreasing this di/dt value, amplitude I RM  is decreased. 
   A problem which arises is that resistor R must be provided to dissipate a reverse overvoltage in inductance L. Indeed, upon the turn-on switching of switch K, the voltage across inductance L takes the value of output voltage V S . The same losses occur at the transistor turning-off. These are resistive losses which are all the greater as the di/dt value is high. In other conventional examples, dissipation element R is replaced with a capacitor, a zener diode, etc. 
   Thus, this second solution has the same disadvantages as the use of a diode with no recovered charges. 
   A third known solution (not shown) consists of a circuit for helping the switching using the transient switching resonance. Such a circuit uses, like the circuit of  FIG. 4 , an additional inductance. However, to avoid resistive loss problems, a second switch, the control of which is desynchronized with respect to that of switch K, is used. 
   An example of a switching aid circuit of this type is described in paper “An overview of soft switching technics for PWM convertors” by G. Hua and F. Lee, published in EPE Journal, Vol. 3, March 1993. 
   Such a solution provides satisfactory results, but has a particularly complex and expensive implementation. In particular, a control system desynchronized from the used switches must be provided. Further, as compared to the circuit of  FIG. 4 , it is necessary to have an additional power switch, two additional diodes and, above all, a high-voltage capacitor. 
   The present invention aims at overcoming the disadvantages of known switching aid circuits. 
   SUMMARY OF THE INVENTION 
   The present invention more specifically aims at providing a switching aid circuit which reduces losses due to the turning-on of a power switch. 
   The present invention also aims at providing a solution requiring no additional switch in a lightly dissipative circuit. 
   The present invention also aims at providing a particularly simple and inexpensive solution. 
   The present invention also aims at providing a solution which is compatible with the use of diodes with recovered charges (PN diodes). 
   The present invention also aims at preserving the control of the di/dt value upon turning-on of the power transistor. 
   To achieve these objects, the present invention provides a circuit for helping the switching of a switched-mode converter, which includes a first inductive power storage element in series with a free wheel diode and a switch, and a second inductive element for controlling the di/dt value upon turning-on of the switch, including: 
   a magnetic circuit having a main winding formed, at least partially, by the first inductive element; 
   means for discharging the second inductive element at the switch turning-off and turning-on; and 
   means for transferring the power corresponding to the turning-on to said main winding. 
   According to an embodiment of the present invention, said discharge means include: 
   a first circuit including a first switching diode; and 
   a second circuit including a first secondary winding of the magnetic circuit. 
   According to an embodiment of the present invention, said transfer means include the first secondary winding of the magnetic circuit and a second switching diode. 
   According to an embodiment of the present invention, the second discharge circuit includes the second inductive element in series with the first secondary winding, the second switching diode, and the switch. 
   According to an embodiment of the present invention, the switching aid circuit further includes a second secondary winding of the magnetic circuit in series with the free wheel diode. 
   According to an embodiment of the present invention, the secondary windings have a same number of turns. 
   According to an embodiment of the present invention, the number of turns of the main winding is greater than the numbers of turns of the secondary windings. 
   The present invention also provides a switched-mode converter of the type including a first inductive power storage element in series with a free wheel diode and a storage element of capacitive type, and a second inductive element for controlling the di/dt value upon turning-on of a switch for cutting-off a supply voltage, including a switching aid circuit. 
   According to an embodiment of the present invention, the converter is of voltage step-up type, the first inductive element forming the main winding of the magnetic circuit being in series with the second inductive element and the switch between two terminals of application of the supply voltage. 
   According to an embodiment of the present invention, the converter is of voltage step-down type, the switch being in series with, among other, the second inductive element and the free wheel diode, between two terminals of application of the supply voltage. 
   The foregoing objects, features and advantages of the present invention will be discussed in detail in the following non-limiting description of specific embodiments in connection with the accompanying drawings, in which: 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1 , previously described, shows a conventional example of a voltage step-up switched-mode converter; 
       FIG. 2 , previously described, shows a conventional example of a voltage step-down switched-mode converter; 
       FIGS. 3A ,  3 B, and  3 C, previously described, illustrate in the form of timing diagrams a problem posed by the circuits of  FIGS. 1 and 2 ; 
       FIG. 4 , previously described, shows another conventional example of a voltage step-up switched-mode converter; 
       FIG. 5  shows an embodiment of a switching aid circuit according to the present invention applied to a voltage step-up converter; 
       FIGS. 6A ,  6 B,  6 C,  6 D,  6 E,  6 F, and  6 G illustrate, in the form of timing diagrams, the operation of the circuit of  FIG. 5 ; 
       FIGS. 7A ,  7 B,  7 C,  7 D,  7 E, and  7 F show the equivalent electric diagrams of the circuit of  FIG. 5  at the different switching phases; and 
       FIG. 8  shows a first embodiment of a switching aid circuit according to the present invention applied to a voltage step-down converter. 
   

   DETAILED DESCRIPTION 
   The same elements have been designated with the same references in the different drawings. For clarity, only those components which are necessary to the understanding of the present invention have been shown in the drawings and will be described hereafter. In particular, the structure of the power switch control circuit has not been detailed and is not part of the present invention, its implementation being within the abilities of those skilled in the art based on the functional indications given in the present description. 
   A feature of the present invention is to provide a magnetic circuit for organizing the discharge of an inductance for controlling the di/dt value, especially, upon closing of the power switch of a switched-mode converter. 
   Another feature of the present invention is to use this magnetic circuit to temporarily store the power generally lost upon switching of the power switch and for storing this power in the converter to the benefit of the load. 
   Another feature of the present invention is to use the inductive element of the circuit for correcting the power factor of the switched-mode converter as an element of the magnetic circuit. 
     FIG. 5  shows the electric diagram of a first embodiment of a voltage step-up switched-mode converter, equipped with a switching aid circuit according to the present invention. 
   As previously, power converter  10  includes a switch K controlled by a circuit (not shown), for example, a pulse-width modulation control circuit (PWM). A power storage inductance L 0  is connected, by a first terminal, to a positive terminal  2  of application of an input voltage V E  provided by a source  6  (for example, a D.C. source). Switch K is in series with an inductance L for controlling the di/dt value, connected to the second terminal  4  of inductance L 0 . The other terminal of switch K is connected to a reference terminal  5  (generally, the ground). Conventionally still, a free wheel diode DL is placed between point  4  and a positive output terminal  3  of the converter. This positive terminal is connected to a first electrode of a storage capacitor C 0  (which may belong to the load Q to be supplied) across which is present output voltage V S . The other terminal of capacitor C 0  is grounded and the anode of diode DL is on the side of terminal  3 . 
   According to the present invention, inductance L 0  belongs to a magnetic circuit  11 , of which it forms the main winding. Magnetic circuit  11  includes two secondary windings L 1  and L 2  having respective numbers of spirals or turns N 1  and N 2  smaller than number N 0  of spirals of inductance L 0 . A first winding L 1  of magnetic circuit  11  is connected in series with diode DL across terminals  3  and  4 . In the example of  FIG. 5 , this inductance has been shown between point  4  and the anode of diode DL. It may also be placed between the cathode of diode DL and terminal  3 , the anode of diode DL being then directly connected to point  4 . A second winding L 2  connects point  4  to terminal  5  by being associated in series with a diode D 2 , the anode of diode D 2  being directed towards ground  5 . As for inductance L 1  and diode D 1 , diode D 2  may be, conversely to what has been shown, connected to point  4 . Finally, a diode D 1  connects point  12  to terminal  3  between inductance L and switch K, the cathode of diode D 1  being connected to point  12 . 
   The function of winding L 1  is, upon turning-off of switch K, to impose a negative voltage across inductance L, to enable it transfer the power that it contains to capacitor C 0 . Diode D 1  is then forward biased. 
   Winding L 2  has the function, upon turning-on of switch K, of imposing a negative voltage across inductance L, to transfer the power that it contains into winding L 2  of the magnetic circuit. This power is recovered by winding L 0  which gives it back to capacitor C 0  at the next switch turning-off. 
   To respect these functionalities, the respective phase points of the windings are chosen as follows. Assuming that the phase point of winding L 0  is connected to terminal  2  as illustrated in  FIG. 5 , the phase point of winding L 1  must be on the side of point  4  and the phase point of winding L 2  must be on the side of ground  5 . Conversely, if the phase point of winding L 0  is connected to point  4 , the phase point of winding L 1  must be on the side of terminal  3  and the phase point of winding L 2  must be on the side of point  4 . 
   The operation of the switching aid circuit shown in  FIG. 5  will be described hereafter in relation with  FIGS. 6A to 6G  and  7 A to  7 F.  FIGS. 6A to 6G  show, in the form of timing diagrams with no scale consideration, an example of a switching cycle of switch K.  FIGS. 7A to 7F  show the equivalent electric diagrams of the circuit of  FIG. 5  in the different switching phases. 
     FIG. 6A  shows voltage V DL  across free wheel diode DL.  FIG. 6B  shows current I DL  in diode DL.  FIG. 6C  shows voltage V K  across switch  4 .  FIG. 6D  shows current I K  in the switch.  FIG. 6E  shows voltage V L  across di/dt-control inductance L.  FIG. 6F  shows current I D1  in diode D 1 .  FIG. 6G  shows current I D2  in diode D 2 . The signs of the currents and voltages shown in  FIGS. 6A to 6G  are taken in relation with the directions indicated in  FIG. 5 . In  FIGS. 7A to 7F , the current flows have been symbolized by arrows. 
   It is assumed that before a time t 10 , switch K is off, the converter then being in free wheel (phase A). During this free wheel period, a current I 0  assumed to be constant flows through diode DL, being given back by inductances L 0  and L 1 . During this phase A where switch K is off, the equivalent diagram of the converter ( FIG. 7A ) only includes inductance L 0  in series with inductance L 1  and diode DL between terminals  2  and  3  to provide the power to the load and to capacitor C 0 . In  FIG. 7A , forward-biased diode DL has been symbolized by a short-circuit. Voltage V DL  across this diode is slightly positive and corresponds to the voltage drop in the forward PN junction (on the order of 0.7 V). Switch K sees across its terminals a voltage V 0  corresponding to voltage V S  plus voltage V DL  and decreased by the voltage drop in winding L 1 . Voltage V L  in inductance L is indeed zero during this period, as will be seen hereafter in relation with the end of the timing diagrams. Diodes D 1  and D 2  are blocked and the currents flowing therethrough are accordingly null. Current I K  in off switch K is of course null. 
   At time t 10 , the turning-on of switch K is controlled. This thus starts a turn-on beginning phase B, the equivalent diagram of which is shown in  FIG. 7B . As compared to  FIG. 7A , the only difference is that inductance L in series with on switch K (short-circuit) is interposed between point  4  and ground  5 . The di/dt value upon turning-on of switch K essentially depends on inductance L. Indeed, this di/dt value depends on voltage V S , on voltage V E , on the mutual inductance of the magnetic circuit and on the off-load inductances L 11  and L 22  of the transformer formed by primary winding L 0 , and secondary windings L 1  and L 2 . Due to the chosen spiral ratio, value L 11  is very large as compared to value L 22 . The mutual inductance is moreover small as compared to value L 11 . As a result, slope (di/dt) is, as a first approximation, equal to V S /L. Current I DL  through diode DL thus decreases with this slope until a time t 12 . Since a PN junction is used, the diode exhibits a recovered charge area. Accordingly, current I DL  annuls at a time t 11 , intermediary between times t 10  and t 12 . Time t 11  corresponds to the time when the current in switch K reaches value I 0 . Between times t 10  and t 12 , diodes D 1  and D 2  remain blocked. Voltage V L  across inductance L becomes approximately equal to voltage V S . 
   At time t 12 , the current through diode DL reaches value I RM  corresponding to the maximum recovered charges. From time t 12 , the charges recovered by diode DL decrease. Diode DL then behaves as a capacitor. The equivalent diagram of this operating phase C is shown in  FIG. 7C  where diode DL has been symbolized in the form of a capacitor. The rest of the elements are the same as in  FIG. 7B . Since the number of spirals of inductance L 1  is small as compared to the number of spirals of inductance L 0 , voltage V L1  thereacross is small. As a result, the capacitance formed by diode DL charges negatively. This phenomenon is illustrated in  FIG. 6B  by a pursuit of the decrease of current I DL  until a time t 13  in the form of a capacitor charge. The current decreases to a current I r  conditioned by inductance L 2 . Indeed, voltage V L , which decreases during this phase C, becomes negative until diode D 2  is turned on when voltage V L  becomes sufficiently negative (time t 13 ). As for diode DL, voltage V DL  reaches, at time t 13 , value −(V S +V L1 +V L2 +V D2 ). Voltage V L  reaches, at time t 13 , value −(V K +V L2 +V D2 ). 
   At time t 13  when diode D 2  turns on, current I DL  through diode DL abruptly stops and the corresponding current is injected back into inductance L 2 . The excess current (I r ) gives the maximum amplitude of the current in inductance L 2 . This current depends on the numbers of spirals N 0  and N 2  of inductances L 0  and L 2 . From time t 13 , diode D 2  conducts (phase D). The equivalent diagram is illustrated in  FIG. 7D . Since diode DL is blocked (non-conducting), capacitor C 0  is disconnected. The magnetic circuit is, during phase D, dissociated from load Q. Diode D 2  is then used as a free wheel element to transfer the power stored by inductance L into the magnetic circuit via winding L 2 . The voltages across diode DL and inductance L remain unchanged. Similarly, switch K being on, the voltage thereacross is zero. Diode D 1  is blocked. When the current is entirely transferred into the magnetic circuit by inductance L 2 , the current therein goes to zero (time t 14 ), which causes a natural blocking of diode D 2 , that is, with a small di/dt. Winding L 2  enables decreasing of the current in switch K by transferring the power to the magnetic circuit which will give it back through inductance L 0 . Between times t 13  and t 14 , the current in switch K will decrease from level I 0 +I r  to level I 0 . 
   At time t 14 , the voltage across inductance L goes to zero, all the power that it contained having been transferred to the magnetic circuit. The voltage across diode DL slightly rises back while remaining negative and takes a value −(V S +V L1 )+V L +V K . It should be reminded that voltages V L  and V K  are negligible (considered as null) with respect to voltages V S  and V L1 . 
   Time t 14  is the beginning of a phase E where the switch is on and where the switching is over. The equivalent diagram is shown in  FIG. 7E . It only includes source  6 , inductances L 0  and L, and switch K. Current I K  is stable at level I 0 , as well as voltage V DL , the free wheel diode being blocked. The voltage across switch K of course is zero, as well as the voltage across inductance L and the currents in diodes D 1  and D 2 . During phase E, inductance L 0  is loaded through inductance L and switch K. 
   At a time t 15  when switch K is turned off, a negative voltage is imposed across inductance L, due to the presence of winding L 1 . It should be noted that, in this case, it is not necessary to control the di/dt value upon turning-off of the transistor (conventionally). The current abruptly stops in switch K. The inversion of the voltage across inductance L 1  causes the discharge, through diode D 1 , of the power stored during phase E in inductance L. At time t 15 , current I D1  thus abruptly takes value I 0  and this current decreases to reach value zero at a time t 16 . The decrease slope of current I D1  is a function of the value of inductance L and approximately corresponds to V L1 /L. The current through inductance L goes to zero at time t 16  and all the current accumulated in winding L 0  then flows through winding L 1  and diode DL. The equivalent diagram of phase F is illustrated in  FIG. 7F . It should be noted that diodes DL and D 1  are on at the same time, but the current through diode DL starts from zero at time t 15 . 
   Time t 16  starts a new phase A where the switch is off. 
   An advantage of the present invention is that it enables recovering the losses due to the turn-on switching of the power switch to inject them back into the load by means of the magnetic circuit. The reinjection of the current into the converter, during turn-on switching phase D of the switch, enables decreasing the duty cycle. The controller (control circuit of switch K) generally automatically decreases this duty cycle by a regulation means which is not part of the present invention. A significant improvement of the converter efficiency is thus here obtained. 
   Another advantage of the present invention is that the provided solution is particularly simple. As compared to the conventional circuit of  FIG. 4 , one power switch and, above all, a complex control circuit, are spared. 
   Another advantage of the present invention is that it requires no modification of the power switch control circuit, provided that said circuit performs (which is generally the case) a regulation. The implementation of the present invention requires adding one magnetic circuit L 0 , L 1 , L 2 , which can be obtained by means of a single three-winding inductance. Such a magnetic circuit is considerably less expensive than the required complexity of the control circuit of  FIG. 4  and than a diode with no recovered charges. On this regard, it should be noted that the solution of a diode with no recovered charges does not enable recovering the losses in the switch. 
     FIG. 8  shows another embodiment of a switching aid circuit  10 ′ of the present invention, applied to a voltage step-down converter. The diagram of  FIG. 8  should be compared to that of  FIG. 2 . As compared to the diagram of  FIG. 2 , inductance L is interposed between point  4 ′ and switch K. Inductance L 2  in series with diode D 2  is connected between terminal  2  and point  4 ′, the anode of diode D 2  being on the side of terminal  2 . Winding L 1  is connected in series with diode DL between point  4 ′ and ground  5 , the anode of diode DL being on the ground side. Finally, diode D 1  connects to ground  5  point  12  between switch K and inductance L, the anode of diode D 1  being grounded. In the example of  FIG. 8 , the phase point of winding L 0  is connected to point  4 ′. Accordingly, to fulfill the described functions of magnetic circuit  11 ′, the phase point of winding L 1  is on the side of ground terminal  5  and the phase point of winding L 2  is on the side of terminal  2 . 
   The operation of the switching aid circuit illustrated in  FIG. 8  can be deduced from the discussion of  FIGS. 5 to 7 . 
   Of course, the present invention is likely to have various alterations, modifications, and improvements which will readily occur to those skilled in the art. In particular, the sizing of the different windings of the magnetic circuit may be modified, provided to respect a winding L 0  having a number of spirals much greater than windings L 1  and L 2 . Preferably, the numbers of spirals of windings L 1  and L 2  are equal, and the number of spirals of winding L 0  is approximately 10 times greater than that of windings N 1  and N 2 . 
   Further, adapting the present invention to a buck-boost converter is within the abilities of those skilled in the art based on the indications given hereabove. 
   Further, the present invention applies to any converter assembly, provided that it is a switched-mode converter. In particular, if in the case of a step-down converter ( FIG. 8 ), the switch has been shown with a terminal connected to the most positive voltage, there also exist assemblies in which this switch has a grounded terminal. The present invention also applies to this type of assembly. It is sufficient to invert the respective positions of series associations K–L and L 1 –DL with respect to point  4 ′, to connect diode D 1  by its cathode to terminal  2 , and to place series association L 2 –D 2  in parallel on association K–L, the cathode of diode D 2  remaining connected to node  4 ′. Inductance L 0  still is connected on the cathode side of free wheel diode DL in series with capacitor C 0 . 
   Finally, among the possible alternatives, inductance L 0  may be divided into a (main) element of the magnetic circuit in series with a distinct inductance that does not belong to the magnetic circuit. The switching speeds of the diodes may also be adapted although, to obtain the advantages of the present invention, these diodes need not be fast. 
   Having thus described at least one illustrative embodiment of the invention, various alterations, modifications, and improvements will readily occur to those skilled in the art. Such alterations, modifications, and improvements are intended to be within and scope of the invention. Accordingly, the foregoing description is by way of example only and is not as limiting. The invention is limited only as defined in the following claims and the equivalents thereto.