Abstract:
A method and system for link adaptation in an orthogonal frequency division multiplexing (OFDM) wireless communication system are disclosed. The entire sub-channels are divided into a plurality of groups. A channel quality indicator (CQI) is generated for each group based on channel quality status in each group, and communication parameters are adjusted in accordance with the CQI.

Description:
CROSS REFERENCE TO RELATED APPLICATION  
       [0001]     This application claims the benefit of U.S. provisional application No. 60/600,741 filed Aug. 11, 2004 which is incorporated by reference as if fully set forth. 
     
    
     FIELD OF INVENTION  
       [0002]     The present invention is related to an orthogonal frequency division multiplexing (OFDM) wireless communication system. More particularly, the present invention is related to a method and system for link adaptation in an OFDM wireless communication system.  
       BACKGROUND  
       [0003]     Current wireless communication systems provide broadband services such as wireless Internet access to subscribers. Those broadband services require reliable and high-rate communications over multi-path fading channels. Orthogonal frequency division multiplexing (OFDM) is one of the solutions to mitigate the effects of multi-path fading. The combination of multiple-input multiple-output (MIMO) and OFDM (OFDM-MIMO) technologies can bring high bandwidth efficiency for local area network (LAN) or wide area network (WAN) environments.  
         [0004]     For an efficient operation of wireless communication systems, a link adaptation for communication parameters is required. Link adaptation is an approach for selecting communication parameters, including a coding rate, a modulation scheme, a transmit power or the like, in order to maximize the throughput.  
         [0005]     In the OFDM-MIMO systems, water-pouring power/bit allocation (WP) is strongly suggested to maximize downlink capacity. In order to determine the WP schemes properly, not only correlation of sub-channels but correlation of sub-channels&#39; power should be known. The transmission of this information requires considerable overhead. Accordingly, it is desirable to have alternate approaches to signaling such information.  
       SUMMARY  
       [0006]     A method and system for link adaptation in an OFDM wireless communication system is provided. The sub-channels are divided into a plurality of groups. A channel quality indicator (CQI) is generated for each group based on channel quality status in each group of sub-channels, and communication parameters on each sub-channel are adjusted in accordance with the CQI. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0007]      FIG. 1  shows the correlation |P k | versus k for several typical values of α when P=256.  
         [0008]      FIG. 2  shows the correlation |P k | versus k for two values of P when α=0.64.  
         [0009]      FIG. 3  shows the correlation γ k  versus k for several typical values of α when P=256.  
         [0010]      FIG. 4  shows the correlation γ k  versus k for two values of P when α=0.64.  
         [0011]      FIG. 5  is a flow diagram of a process for adjusting communication parameters.  
         [0012]      FIG. 6  shows generation of CQI q   (t)  for each group of sub-channels.  
         [0013]      FIG. 7  is a diagram of a system for link adaptation. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0014]     Hereinafter, the following embodiments are explained with reference to IEEE 802.11 system. However, it should be noted that the embodiments are not limited to the IEEE 802.11 system, but may be applicable to any wireless communication system.  
         [0015]     Suppose h (t,r) ={h 0   (t,r) ,h 1   (t,r) , . . . , h W-1   (t,r) } is a time-domain channel response vector of length W for the channel between the tth transmit antenna and the rth receive antenna. The average power of the coefficient h l   (t,r)  is expressed by σ 1   2 =E{|h l   (t,r) | 2 } which is independent of the values of t and r. This is because the size of the antenna array in MIMO systems is usually much less than the propagation distance of the first arrival path.  
         [0016]     In IEEE 802.11 a/n, a 20-MHz sampling rate is used, resulting in a 50-ns time resolution of the channel response. Normalized power-delay profile can be expressed as  
                     σ   l   2     =       G     -   1       ⁢     ⅇ       -   l     /     (     Γ   /   50     )                         =       G     -   1       ⁢     ⅇ       -   α     ·   l           ,                 (     Equation   ⁢           ⁢   1     )             
 
 where α=50/Γ,  
       G   =         ∑     l   =   0       W   -   1       ⁢     ⅇ       -   α     ·   l         =         1   -     ⅇ       -   α     ·   W           1   -     ⅇ     -   α           ≈     1     1   -     ⅇ     -   α                   
 
 for αW&gt;&gt;1, and Γ in nanoseconds is the power-delay time constant for the paths (clusters). 
 
         [0017]     Summing the average power of the coefficients over the delay spread, W, results in  
           ∑     l   =   0       W   -   1       ⁢     σ   l   2       =   1.       
 
 The parameter Γ depends on the propagation distance of the first path (D 0 ) and path loss model of the channel. To evaluate the average power of different paths, the propagation distances of these paths should be known. Because the sampling duration is 50 ns in the foregoing example, the propagation distance between two consecutive paths is 15 meters. Therefore, if D l  denotes the propagation distance of the lth path in meters, D l+1 =D l +15 for l=0, 1, . . . , W−2. Without loss of generality, only the power loss ratio of the second path to the first path may be considered, which is defined as  
               R   loss     =     {             (         D   0     +   15       D   0       )     2             if   ⁢           ⁢     D   0       ≤     D   free                   (         D   0     +   15       D   0       )     3.5             if   ⁢           ⁢     D   0       &gt;     D   free                       (     Equation   ⁢           ⁢   2     )             
 
 where D free  space propagation distance. When D 0 ≦D free , the channel is line-of-sight (LOS). Otherwise, the channel is non-LOS. When R loss  is given, the parameter α and power-delay time constant Γ can be calculated by solving the equation 
 
e α =R loss .  (Equation 3) 
 
         [0018]     Assuming D free =15 m, the values of α and Γ are shown in Table 1 for several typical values of D 0 . The average value of Γ within a room is approximately 60 ns.  
                                     TABLE 1                       D 0  in meters   α   Γ in nanoseconds                                15   1.38   70       45   1.0   50       75   0.64   32       100   0.49   24                  
 
         [0019]     Suppose H (t,r) ={H 0   (t,r) ,H 1   (t,r) , . . . , H P−1   (t,r) } is the frequency-domain channel response vector of length P for the channel between the tth transmit antenna and the rth receive antenna. In other words, H (t,r)  consists of P sub-channels. The Pth sub-channel can be represented as  
                 H   p     (     t   ,   r     )       =       ∑     l   =   0       W   -   1       ⁢       h   l     (     t   ,   r     )       ⁢     W   P     l   ·   p             ,           (     Equation   ⁢           ⁢   4     )             
 
 where W P =e −j2π/P . The correlation between the P 1 th and P 2 th sub-channels is defined as 
 
 P   P1,P2   (t,r)   =E└H   P1   (t,r) ( H   P2   (t,r) )*┘.   (Equation 5) 
 
         [0020]     Suppose h l   (t,r)  is a complex Gaussian variable with zero mean and is independent of h m   (t,r)  if l≠m. According to Equation 1,  
                     ρ       p   1     ,     p   2         (     t   ,   r     )       =     E   ⁡     [       (       ∑     l   =   0       W   -   1       ⁢       h   l     (     t   ,   r     )       ⁢     W   P     l   ·     p   1             )     ·     (       ∑     m   =   0       W   -   1       ⁢         (     h   m     (     t   ,   r     )       )     *     ⁢     W   P     m   ·     p   2             )       ]                     =       G     -   1       ⁢       ∑     l   =   0       W   -   1       ⁢       ⅇ       -   α     ⁢           ⁢   l       ⁢     W   P       (       p   1     -     p   2       )     ⁢   l               ,                 (     Equation   ⁢           ⁢   6     )             
 
 which is independent of the values of t and r. Assuming k=P 1 P 2  for k=0,1, . . . , P- 1  and αW&gt;&gt;1, Equation 6 can be written as  
               ρ   k     =         1   -     ⅇ     -   α           1   -       ⅇ     -   α       ⁢     ⅇ       -   j     ⁢           ⁢   2   ⁢   π   ⁢           ⁢     k   /   P               .             (     Equation   ⁢           ⁢   7     )             
 
 The variable k represents the number of sub-channels spaced between the two sub-channels under consideration. From Equation 7,  
               |     ρ   k     |     =         1   -     ⅇ     -   α             1   -     2   ⁢     ⅇ     -   α       ⁢     cos   ⁡     (     2   ⁢   π   ⁢           ⁢     k   /   P       )         +     ⅇ       -   2     ⁢           ⁢   α             .             (     Equation   ⁢           ⁢   8     )             
 
         [0021]      FIG. 1  shows the curves of |P k | against k for several typical values of α when P=256. With the decrease of the parameter α, the correlation between the two sub-channels spaced with k sub-carriers is reduced. According to Equation 1, the smaller the parameter α, the more comparable the average power of the paths. In other words, such a channel consists of more effective multi-paths and therefore the channel becomes more frequency-selective. In the limit case that a α→0,|P k |→0 for any value of k. On the other hand, if the channel is flat fading (non frequency-selective), α→∞, resulting in |P k |=1 for all values of k.  
         [0022]      FIG. 2  shows the curves of |P k | versus k for a different number of sub-channels P when α=0.64. With the decrease of P, the correlation curve becomes narrow linearly. For example, the sub-channels with |P k |≧0.9 for P=64 and P=256 have to be spaced less than 4 and 16 sub-carriers, respectively.  
         [0023]     In order to use the principle of “water-filling”, a measure for CQI must be defined. The CQI should be constructed based on the power of the sub-channels. Although |P k | presents the correlation between two sub-channels spaced by k sub-carriers, it does not show clearly the correlation of the two sub-channels&#39; power. Therefore, the correlation of sub-channels&#39; power should be derived. The correlation of sub-channels&#39; power is defined as  
               γ       p   1     ,     p   2         (     t   ,   r     )       =         E   ⁡     [     |     H     p   1       (     t   ,   r     )       ⁢     |   2     |     H     p   2       (     t   ,   r     )       ⁢     |   2       ]         E   ⁡     [     |     H     p   1       (     t   ,   r     )       ⁢     |   4       ]         .             (     Equation   ⁢           ⁢   9     )             
 
         [0024]     With αW&gt;&gt;1,  
                 E   ⁡     [     |     H     p   1       (     t   ,   r     )       ⁢     |   4       ]       =     2   -       1   -     ⅇ     -   α           1   +     ⅇ     -   α               ,           (     Equation   ⁢           ⁢   10     )             
 
 and  
                     E   ⁢           [              H     p   1       (     t   ,   r     )            2     ⁢            H     p   2       (     t   ,   r     )            2       ]     =       ⁢     1   -       1   -     ⅇ     -   α           1   +     ⅇ     -   α           +                     ⁢           (     1   -     ⅇ     -   α         )     2       1   -     2   ⁢     ⅇ     -   α       ⁢   cos   ⁢           ⁢     (     2   ⁢   π   ⁢           ⁢     k   /   P       )       +     ⅇ       -   2     ⁢   α           ,                   (     Equation   ⁢           ⁢   11     )             
 
 where k=P 1 -P 2  ε[0,W−1]. Equation 10 and Equation 11 are independent of the values of t and r. In the derivation of Equation 10 and Equation 11, it is assumed that the real and imaginary parts of a multi-path coefficient, (say h P   (t,r)  for P ε[0,W-1]), have the same variance and are independent from each other. Substitution of Equation 10 and Equation 11 into Equation 9 results in  
               γ   k     =                 2   ⁢       ⅇ     -   α       ⁡     [     1   -     2   ⁢     ⅇ     -   α       ⁢   cos   ⁢           ⁢     (     2   ⁢   π   ⁢           ⁢     k   /   P       )       +     ⅇ       -   2     ⁢   α         ]         +                   (     1   -     ⅇ     -   α         )     2     ⁢     (     1   +     ⅇ     -   α         )                 (     1   +     3   ⁢     ⅇ     -   α           )     ⁢     (     1   -     2   ⁢     ⅇ     -   α       ⁢   cos   ⁢           ⁢     (     2   ⁢   π   ⁢           ⁢     k   /   P       )       +     ⅇ       -   2     ⁢   α         )         .             (     Equation   ⁢           ⁢   12     )             
 
         [0025]      FIG. 3  shows the curves of γ k  against k for several typical values of α when P=256. From  FIG. 3 , the smallest value of the correlation γ k  is around 0.5 at k=P/2. In other words, two sub-channels spaced with P/2 sub-carriers may statistically have about 3 dB differences in power. Therefore, it is not necessary to report the CQI for each of the sub-channels.  FIG. 4  shows the curves of γ k  versus k for a different number of sub-carriers P when α=0.64. The curves are shrunk linearly as the value of P is reduced.  
         [0026]      FIG. 5  is a flow diagram of a process  500  for link adaptation in accordance with the present invention. Sub-channels are divided into a plurality of groups (step  502 ).  FIG. 6  shows a scheme for generating the CQI in each group of sub-channels. In  FIG. 6 , the total sub-channels are divided into Q groups and each group consists of Δ consecutive sub-channels with Δ=P/Q. The correlation of the sub-channels&#39; power in a group for different values of Q is shown in Table 2.  
                                     TABLE 2                               Statistical differences in               power between two sub-       The values of Q   γ k  for 0 ≦ k ≦ Δ − 1   channels in a group                                20   ≧0.9   0.46 dB       16   ≧0.8   0.97 dB       8   ≧0.6   2.22 dB                    
         [0027]     A CQI is generated for each group based on channel quality status in each group (step  504 ). The channel quality status may be analyzed by different methods including, but not limited to, a signal-to-noise ratio (SNR), a bit error rate (BER), a packet error rate (PER), or the like. Hereinafter, the following embodiment is explained with reference to an SNR. However, it should be understood that other methods may be implemented alternatively. Assuming that CQI q   (t)  denotes the qth CQI of the tth transmit antenna (q=0,1, . . . , Q-1 and t=0,1, . . . , N r −1), CQI q   (t)  is preferably calculated as 
 
 CQI   q   (t)   =B+└ 10 log 10 ( SNR   q   (t) )┘,   (Equation 13) 
 
 where └x┘ is the largest integer smaller or equal to x, B is an integer which should be determined based on system requirements. SNR is calculated as  
               SNR   q     (   t   )       =       1     σ   2       ⁢       ∑     r   =   0         N   R     -   1       ⁢           ⁢       ∑     l   =   0       Δ   -   1       ⁢           ⁢              H     l   +     q   ⁢           ⁢   Δ         (     t   ,   r     )            2     .                   (     Equation   ⁢           ⁢   14     )             
 
 N R  is the number of receive antennas and σ 2  is the noise variance in each sub-channel. 
 
         [0028]     The CQI is fed back to adjust communication parameters (step  506 ). Since CQI is generated based on the sub-channels in a group, total number of Q×N T  CQIs are generated in a transmission frame (packet), where N T  is the number of transmit antennas. It is not necessary to report CQI on an OFDM symbol basis, since the channel may change little in a frame (packet) interval; and due to common phase error (CPE) invoked by the combination of RF oscillator and the phase-locked loop, the phase of the channel responses may change. However, such a change does not affect the power of the sub-channels. Therefore, the CQI can be calculated based on the channel responses estimated from the long training sequences on a frame basis without using the pilot tones inserted in OFDM symbols. The inserted pilot tones are used only for the purpose of correcting the CPE.  
         [0029]     For example, if each of the CQI indicates one of four states that correspond to the modulation schemes (BPSK, QPSK, 16QAM, 64 QAM), a number of 2×Q×N T  bits are required to report all of the CQIs. In a typical case that Q=16 and N T =4, 2×Q×N r =128 bits are required to report the CQIs. This is reasonable as compared to the number of data in a transmission frame. Alternatively, the CQI may represent a combination of two or more communication parameters, such as a combination of a coding rate and a modulation order.  
         [0030]     Because any pair of sub-channels statistically has a maximum of 3 dB differences in power, the CQI reported according to Equation 13 may be more meaningful for the change of coding rates rather than modulation schemes. Therefore, the modulation scheme may be kept constant for all the sub-channels while adjusting the coding rate according to the reported CQI for different groups of the sub-channels. In this case, the modulation scheme may be determined according to 
 
 M   (t)   =C+└ 10 log 10 ( SNR   (t) )┘,   (Equation 15) 
 
 where C is an integer which should be determined based on system requirements. SNR is determined as follows:  
               SNR     (   t   )       =         1     σ   2       ⁢       ∑     r   =   0         N   R     -   1       ⁢            H   0     (     t   ,   r     )            2         =       1     N   R       ⁢       ∑     r   =   0         N   R     -   1       ⁢           ⁢       ∑     l   =   0       W   -   1       ⁢           ⁢              h   l     (     t   ,   r     )            2     .                     (     Equation   ⁢           ⁢   16     )             
 
         [0031]     Optionally, after channel estimation, paths with relatively strong power may be selected. After the selection of the paths having relatively strong power, the number of effective paths is reduced to M that is usually less than W. Suppose  
           G     m   =       (   t   )       ⁢       ∑     r   =   0         N   R     -   1       ⁢       h   m     (     t   ,   r     )       ⁢           ⁢   for   ⁢           ⁢   m         ∈     [     0   ,     M   -   1       ]         
 
 is the effective channel response and K is the vector indicating the locations of the M paths. With G m   (t)  and K, first all the sub-channels of each antenna can be calculated using Equation 3 and then the modulation and coding schemes can be decided for optimization. Optionally the MIMO channel matrix of a reference sub-carrier may be transmitted so that calibration can be made. 
 
         [0032]     Some embodiments for selecting and indexing the reference subcarriers are as follows. In one embodiment, the network configures the reference subcarriers and the index of the subcarrier(s) are known to both the network and the subscriber. Accordingly, typically, the index of the reference subcarier(s) is not reported to the transmitter. In another embodiment, the receiver can dynamically choose reference subcarriers based on instantaneous channel transfer functions of all subcarriers and other factors in the spectrum. The receiver chooses the index of the reference subcarrier and reports the index to the transmitter.  
         [0033]      FIG. 7  is a diagram of a system  700  for link adaptation. The system  700  comprises a CQI generator  702  and a link adaptor  704 . The CQI generator  702  generates a CQI based on channel quality status of received signals  706  via each group of sub-channels. A CQI  708  generated by the CQI generator  702  is forwarded to the link adaptor  704  for generating control signals  710  for adjusting communication parameters. The communication parameters include, but are not limited to, a coding rate, a modulation mode, a transmit power level or the like. The link adaptor  704  may comprise a look-up table for adjusting communication parameters in accordance with the input CQI. The CQI generator  702  may reside at a wireless transmit/receive unit (WTRU), base station or both. The link adapter may reside at a WTRU, base station or both.  
         [0034]     The MIMO-OFDM transmitter and/or receiver of the above embodiments may be used in a WTRU or base station. The transmitter and/or receiver elements may be implemented as a single integrated circuit (IC), multiple ICs, logical programmable gate array (LPGA), discrete components or a combination of any of these IC(s), LPGA, and/or discrete components.  
         [0035]     A WTRU includes but is not limited to a user equipment, mobile station, fixed or mobile subscriber unit, pager, or any other type of device capable of operating in a wireless environment. A base station includes but is not limited to a Node-B, site controller, access point or any other type of interfacing device in a wireless environment.  
         [0036]     Although the features and elements of the present invention are described in the preferred embodiments in particular combinations, each feature or element can be used alone without the other features and elements of the preferred embodiments or in various combinations with or without other features and elements of the present invention.