Abstract:
The current invention relates, inter alia, to charge pulse amplitude and time detecting circuits, offering very low amplitude and temporal noise, and overcoming noise performance limits in charge pulse detection circuits according to prior art. Embodiments of the present invention may include a sensing device delivering charge pulses onto a sense node, an active buffer buffering the voltage on the sense node with a low impedance, a recharge device removing signal charge from the sense node, a noise filter connected to the output of the active buffer transmitting signal voltage pulses while attenuating noise from the recharge device. Additional and alternative embodiments are specified and claimed.

Description:
FIELD OF THE INVENTION 
       [0001]    The current invention generally relates to charge pulse and current pulse amplitude and time detecting circuits. In particular, the invention relates to charge pulse detecting circuits using optoelectronic sensing devices as well as arrays thereof and to X-ray photon detecting and counting applications. 
     
    
     
       DESCRIPTION OF THE FIGURES 
         [0002]    Features and advantages of the invention will become apparent in the light of the ensuing description of some embodiments thereof, given by way of example only, with reference to the accompanying figures, wherein: 
           [0003]      FIG. 1  is an schematic illustration of a charge pulse detecting circuit according to the prior art; 
           [0004]      FIG. 2  is a schematic illustration of an embodiment of a general architecture of a charge pulse detecting circuit according to an embodiment of the invention; 
           [0005]      FIG. 3  is a schematic illustration of a particular charge pulse detecting circuit employing MOS transistors, according to an embodiment of the invention; 
           [0006]      FIG. 4  is an schematic illustration of an input signal current to output voltage transimpedance function that corresponds to the embodiment schematically illustrated in  FIG. 3 ; 
           [0007]      FIG. 5  is a schematic illustration of the unfiltered and filtered power spectral densities of the noise generated by a recharge device corresponding to the embodiment schematically shown in  FIG. 3 ; 
           [0008]      FIG. 6  is a schematic illustration of a particular charge pulse detecting circuit comprising an inverting voltage amplifier and a recharge device connected between a sense node and the output of the inverting voltage amplifier, according to an alternative embodiment of the invention; 
           [0009]      FIG. 7  is a schematic illustration of a particular charge pulse detecting circuit, comprising an active band-pass type noise filter imparting voltage amplification to the signal voltage pulses, according to another embodiment of the invention; and 
           [0010]      FIG. 8  is a schematic illustration of a particular charge pulse detecting circuit wherein the recharge device is embodied by a reset switch that is closed for resetting the sense node and left open during pulse detection, according to a yet alternative embodiment of the invention. 
       
    
    
     LIST OF ABBREVIATIONS 
       [0000]    
       
         RMS root mean square 
         DC direct current 
         AC alternating current 
         MOS metal oxide semiconductor 
         CMOS complementary metal oxide semiconductor 
         PSD power spectral density 
         W/L ratio of the gate width over the gate length of a MOS transistor 
       
     
       BACKGROUND OF THE INVENTION 
       [0018]    Current pulse detecting circuits are used for a wide range of applications including sensors which contain a sensing device able to deliver electrical charge representing the sensed physical property. In order to detect minute changes of the sensed physical property, detecting circuits providing a high charge to voltage conversion factor and low noise at high bandwidth are of a major interest. 
         [0019]    With reference to  FIG. 1 , state of the art charge pulse detecting circuits usually comprise a sensing device  101  delivering an amount of charge which represents the sensed physical property to an input node  111 , an inverting amplifier  102  and a sense capacitor  103  configured to form a capacitance feedback amplifier, a recharge resistor  104  in parallel with sense capacitor  103  and an input capacitor  105  which may be a parasitic capacitance. For short current pulses delivered by sensing device  101  and high values of recharge resistor  104 , amplifier  102  produces on an output node  112  a voltage pulse with a pulse height defined by the integrated charge of the input current pulse and the capacitance value of sense capacitor  103 . The input charge is subsequently slowly removed from input node  111  across recharge resistor  104 , and a stable DC operation voltage point is established on input node  111  by feedback operation of amplifier  102  and recharge resistor  104 . State of the art charge pulse detecting circuits are described in G. Lutz, “Semiconductor Radiation Detectors”, pp. 190, Springer, Berlin; Heidelberg. 
         [0020]    Circuit analysis shows that, for the case of an inverting amplifier  102  with a sufficiently high gain-times-bandwidth product, the charge detecting circuit&#39;s input current to output voltage transimpedance function may be approximated as: 
         [0000]    
       
         
           
             
               
                 
                   
                     
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         [0000]    wherein v o  is the ac voltage on output node  112 , i in  is the ac current delivered by sensing device  101 , C l  is a load capacitance connected to output node  112 , gm A  is the transconductance of amplifier  102 , R r  is the resistance value of recharge resistor  104 , C s  is the capacitance value of sense capacitor  103 , C i  is the sum of capacitance from input node  111  to any ac ground node, and s is the complex signal frequency. Note that alternative mathematical terms may be used to represent the approximation of the transimpedance function. 
         [0021]    For frequencies above ½πR r C s  but below the zero frequency and the second pole frequency, the transimpedance is approximately equal to 1/sC 5  i.e. to the sense capacitance impedance. Therefore, a high charge to voltage conversion factor may be achieved if C s  is small and R r C s  is longer than the width of the detected current pulses. 
         [0022]    The major noise sources in the discussed state of the art current pulse detecting circuit are amplifier  102  and recharge resistor  104 . Noise contributed by amplifier  102  can be arbitrarily reduced by increasing load capacitance, amplifier transconductance and amplifier transistor device area. Circuit analysis under the same assumptions as above yields the following approximation of the output noise power spectral density due to the thermal noise caused by recharge resistor  104 : 
         [0000]    
       
         
           
             
               
                 
                   
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         [0023]    Note that alternative mathematical expressions may be used to represent the approximation of the recharge resistor noise power spectral density. Neglecting the effects of the zero and second pole at high frequency, the input node charge RMS variation q ni, Rr, prior art  equivalent of the recharge resistor noise can be approximated as: 
         [0000]    
       
         
           
             
               
                 
                   
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                       ni 
                       , 
                       
                           
                       
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         [0000]    wherein k is the Boltzmann Constant and T is the absolute temperature in K. Note that different alternative expressions may be used to represent the recharge resistor noise equivalent input charge. We note that low capacitance values of sense capacitor  103  are desirable in order to achieve low detecting circuit noise i.e. high signal to noise ratio at low charge amounts delivered by sensing device  101 . There are, however, practical limits to the capacitance value Cs, mainly determined by the constraints of fabrication processes. Therefore, the noise performance of state of the art pulse detecting circuits is limited for a given process technology. 
       DESCRIPTION OF THE INVENTION 
       [0024]    The present invention refers, inter alia, to charge pulse detecting circuits. An embodiment of the present invention may comprise a sensing device delivering charge pulses representing the amplitude of a sensed physical property onto a sense node, an active buffer circuit buffering a voltage from the sense node to a buffer output node, a recharge device recharging the sense node to a physical or virtual DC potential with a high but finite DC impedance and a noise filter circuit filtering the output voltage of the buffer, to reduce noise from the recharge device and providing an output voltage on an output node. The noise due to the recharge device observed on said output node is reduced to an equivalent RMS variation of, for example, less than 50, 40, 30 or 20 electrons or holes. This is, for example, several times less than the value of the unfiltered noise due to the recharge device observed on the sense node. For example, the filtered noise may be, for example, 0.5, 0.3, 0.2 or 0.1 times the value of the unfiltered noise. 
         [0025]    Embodiments may use employ a MOS source follower circuit as an active buffer. 
         [0026]    Embodiments may employ, as an active buffer, an inverting voltage amplifier providing a voltage gain greater than unity. 
         [0027]    In embodiments the recharge device may be embodied by a resistor connected between the sense node and a DC potential. 
         [0028]    In embodiments the recharge device may be embodied by a recharge resistor connected between the sense node and the output node of the inverting voltage amplifier with its input node connected to the sense node as well. 
         [0029]    Embodiments may employ, as a recharge device, a switch that connects the sense node to a the physical or virtual DC potential with low impedance in first state, which may be established in regular or irregular intervals of time, and that, in a second state, isolates the sense node from said DC potential through a relatively high, ideally virtually infinite impedance in connection with this embodiment. It should be noted that the term “isolate” and grammatical variations thereof also encompasses the meaning “substantially” isolate. If a MOS transistor is used as a switch, the impedance of said switch may be, for example, higher than 10 12  Ohms in its open state. 
         [0030]    Embodiments may use a noise filter providing band-pass or high-pass frequency domain behaviour. The noise filter may incorporate an active circuit yielding a charge pulse detecting circuit with an output impedance low enough to drive load capacitors of, for example, up to several pico-Farads. 
         [0031]    Embodiments may use, as a noise filter, an active band-pass or high-pass filter providing voltage amplification in its passing band of frequencies. 
         [0032]    Embodiments of the invention allow asynchronous continuous amplitude and arrival time detection of charge pulses with a very low detection limit ranging, for instance, from two to 100 electrons or holes per pulse only. The invention exploits knowledge of the width of the detected pulses through noise filtering in unused frequency ranges. Embodiments of the invention may be adapted to rather long pulses, e.g. in the range of 5, 3, 4 or 1 microseconds, as well as for quite short pulses, e.g. in the range of 20, 10, 5, 3 or 1 nanosecond, using standard integrated circuit fabrication technologies. Embodiments of the invention can be built as compact circuits that may be used as pixel circuits in one-dimensional or two-Dimensional integrated circuit sensor arrays. 
         [0033]    In the following, a description of some embodiments of the present invention is provided. These embodiments should be considered as examples and their choice is not to be construed as limiting. Modifications of the described embodiments may be apparent to those skilled in the art without deviating from the scope of the invention. 
         [0034]    Referring to  FIG. 2  an embodiment of the invention may possibly but not necessarily comprise the following elements:
       a sensing device  201  delivering providing to a sense node  211  a charge pulse representing the magnitude of the a sensed physical property.   an active buffer  202  with its input connected to sense node  211  and its output connected to a buffer output node  212 . In one embodiment, active Buffer  202  may provide voltage amplification. In another embodiment, active buffer  202  does not provide voltage amplification.   a recharge device  204  with at least one first terminal connected to sense node  211  and at least one second terminal connected to either a fixed potential or buffer output node  212 .       
 
         [0038]    a continuous time noise filter  206  with its input connected to buffer output node  212  and its output connected to an output node  213 .
       a sense capacitor  205  connected between sense node  211  and a DC potential. Sense Capacitor  205  may be a parasitic capacitance.       
 
         [0040]    A charge pulse delivered provided by sensing device  201  onto sense node  211  results in a transition of the voltage on sense node  211  with a transition time substantially equal to the charge pulse width. Please note that the term “equal” as used herein also encompasses the meaning “substantially equal”. In the case of negative signal charge, for instance, a falling voltage edge is obtained. This voltage transition is hereinafter referred to as “signal edge” in the text hereinafter. 
         [0041]    Recharge device  204  subsequently removes the signal charge from sense node  211  and establishes a well defined DC voltage on sense node  211 . During the described recharge process we obtain a voltage transition on sense node  211  opposite to the signal edge. This voltage transition is hereinafter referred to as “recharge transition”. Recharge device  204  is designed such that the recharge time, i.e. the duration of the recharge transition, is significantly longer, e.g. at least twice as long, than the duration of the signal edge, i.e. the charge pulse width. Implementations of the recharge device therefore include but are not limited to high but finite DC impedance paths to a fixed voltage or high but finite DC impedance paths to buffer output node  212  in case active buffer  202  is an inverting voltage amplifier. In the latter case a stable DC input voltage is established by feedback operation of active buffer  202  and recharge device  204 . It should be noted that the term “stable” as used herein also encompasses the term “substantially” stable. 
         [0042]    It is worth mentioning that in correspondence to the relatively long recharge time, the noise bandwidth of the voltage noise power spectrum on sense node  211  is quite small, and the voltage noise power spectral density on sense node  211  is relatively high due to the high DC impedance of recharge device  204 . 
         [0043]    Active buffer  202  is used in order to provide a voltage signal, representing the sense node voltage, driven with low impedance while keeping the impedance of sense node  211  high i.e. the capacitance value of sense capacitor  205  low. Active buffer  202  may or may not provide voltage gain and may be inverting or non-inverting. Note that both the signal edge as well as the recharge transition are reproduced on buffer output node  212 . 
         [0044]    Noise filter  206  generally is a continuous time filter eliminating noise from unused frequency ranges while transmitting the signal edge. It should be noted that the term “eliminating” also encompasses the term “substantially eliminating”. In particular, low frequencies, where most of the recharge device noise power resides, are filtered out. This may involve filtering of the recharge transition. 
         [0045]    Noise filter  206  may be a high pass-filter or a band-pass filter. Note that a high-pass filter commonly generates large high-frequency noise itself. A band-pass filter limiting the bandwidth of its self-generated noise commonly contributes less self-generated noise and may thus be preferable. When using a band-pass filter, however, particular attention has to be paid to the choice and control of the upper band limit frequency, in order to avoid undesired attenuation of the signal edge. 
         [0046]    Noise filter  206  may be passive or active and may or may not apply a voltage gain greater than unity to the signal edge. 
         [0047]    Sensing devices with charge output include but are definitely not limited to optoelectrical sensors such as, for example, homojunction photodiodes, heterojunction photodiodes, pinned-photodiodes and/or photogate type detectors, as employed for example in Charge Coupled Devices (CCDs). 
         [0048]    Reference is now made to  FIG. 3 , which schematically illustrates an embodiment of the present invention using MOS transistors where recharge device  204  is implemented as recharge resistor  304  with at least one first terminal connected to sense node  311  and at least one second terminal connected to a DC potential. 
         [0049]    Active buffer  202  is implemented as a unity gain source follower buffer comprising a source follower transistor  302  and a current source transistor  303 . 
         [0050]    Noise filter  206  comprises a high-pass filter capacitance  306  and a high pass filter resistor  307  forming a passive high-pass filter, as well as a band-limiting source follower transistor  308 , a current source transistor  309  and a band-limiting capacitor  310  forming an active low-pass filter cascaded with the passive high-pass filter. The described configuration results in an actively buffered band-pass filter with unity gain in the passing band of frequencies, with its input being a buffer output node  312  and its output being the output node  313  of the pulse detecting circuit. 
         [0051]    As explained above, the resistance of recharge resistor  304  needs to be relatively high in order to limit the bandwidth of the recharge resistor noise to relatively low frequencies. For a charge pulse detecting circuit able to detect pulses of relatively high width, for example up to a microsecond, with a capacitance value of sense capacitor  305  of, for instance, 2 to 20 femto-Farads, the resistance value of recharge resistor  304  needs to be in the range of, for instance, 10 9  Ohms in order to limit the noise of recharge resistor  304  to sufficiently low frequencies. 
         [0052]    We also mentioned before that noise filter  206  needs to transmit the signal edge; this demands that the time constant of the high pass filter, which may comprise high-pass filter capacitor  306  and high-pass filter resistor  307 , needs to be larger than or equal to the width of the detected charge pulses. For a charge pulse detecting circuit that can be used in 2-dimensional arrays manufactured using typical semiconductor processing technology the capacitance value of high-pass filter capacitor  306  is practically limited to values, for example, below 2 pico-Farad, 1 pico-Farad, or 0.5 pico-Farad affecting a corresponding impedance, in order to avoid excessive circuit area. If such a charge pulse detecting circuit is used to detect charge pulses of a width in the range of, for instance, a microsecond, the required resistance value of high-pass filter resistor  307  ranges, for instance, from 1 Mega-Ohm to several tens of Mega-Ohms such as 20, 30, 40, 50, 60, 70, 80 or 90 Megaohms. 
         [0053]    Implementations of recharge resistor  304  and high-pass filter resistor  307  providing the required high resistance using MOS processing technology may include, inter alia, MOS transistors operated in strongly inverted non-saturated region (triode region) and MOS transistors operated with a weakly inverted channel (sub-threshold operation) and low drain-source voltage. 
         [0054]    The discussed examples of resistor implementations should not be construed as limiting. Different implementations will be apparent to those skilled in the art without deviating from the scope of the present invention. Furthermore it should be noted that different transistor types than depicted in  FIG. 3  may be used in order to achieve equivalent functionality. 
         [0055]    Referring now to  FIG. 4 , the amplitude of the signal transimpedance function  401  of the pulse detecting circuit depicted in  FIG. 3  gives the ratio of the ac voltage v out  on output node  313  to an ac input current i in  provided to sense node  311  by sensing device  301 . 
         [0056]    For frequencies f in the passing band of the band-pass filter, i.e. between the high-pass filter transition frequency  403  approximately equal to ½πR hp C hp  and the band limiter transition frequency  404  approximately equal to gm 2 /2πC l , the signal transimpedance function substantially corresponds to 1/sC s , i.e. the signal charge to voltage conversion factor is defined by the sense capacitor  305  and is essentially constant. In the expressions mentioned above, R hp  is the resistance of high-filter resistor  307 , C hp  is the capacitance of high-pass filter capacitor  306 , gm 2  is the transconductance of band-limiting source follower transistor  308  and C l  is the capacitance of band-limiting capacitor  310 . 
         [0057]    Referring now to  FIG. 5 , the unfiltered recharge resistor noise PSD  501  corresponds to the noise power spectral density of recharge resistor  304  observed on sense node  311  which is essentially described by a low-pass function with a DC PSD level  504  at a value of 4kTRr and its transition frequency corresponding to recharge transition frequency  402 . Note that high resistance values of recharge resistor  304  lead to unfiltered recharge noise PSD  501  with a high DC PSD and a low bandwidth. The statistical charge RMS variation q n,sN , equivalent to the voltage noise on sense node  311 , is found to be: 
         [0000]      q n,SN =√{square root over (kTC s )}  (4)
 
         [0058]    The filtered recharge resistor noise PSD  502  corresponds to the noise power spectral density observed on output node  313  caused by recharge resistor  304  computed under the approximation of exact unity gain for both the active buffer consisting of source follower transistor  302  and current source transistor  303 , as well as the band-limiting buffer comprising band-limiting source follower transistor  308  and current source transistor  309 . Note that filtered recharge resistor noise PSD  502  has a constant maximum PSD level  505  corresponding to 4kT R hp   2 C hp   2 /R r C s   2  and a pole takes effect at high-pass filter transition frequency  403 . The RMS output noise due to recharge resistor  304  corresponds to the square root of the integral of filtered recharge resistor noise PSD  502  over the entire frequency range. Approximation  503  of filtered recharge resistor noise PSD  502  is an approximation that results in an overestimated but simple expression for the RMS output noise. Using approximation  503  we find the following expression term of the input charge RMS variation equivalent to the noise of recharge resistor  304 : 
         [0000]    
       
         
           
             
               
                 
                   
                     q 
                     
                       ni 
                       , 
                       
                           
                       
                        
                       Rr 
                     
                   
                   ≤ 
                   
                     
                       kT 
                        
                       
                         
                           
                             R 
                             hp 
                           
                            
                           
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                           r 
                         
                       
                     
                   
                 
               
               
                 
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         [0000]    Comparing this result to EQN. 4 we find that the noise filter attenuates RMS noise from the recharge resistor  304  with a factor of the square root of the ratio of the recharge transition frequency  402  over the high-pass filter transition frequency  403 . Consequently, the noise reduction factor can be approximated by the term: √{square root over (R hp C hp /R r C s )}. 
         [0059]    Note that, despite the fact that according to the present invention the input charge RMS variation equivalent to the recharge resistor noise is substantially independent of the capacitance of sense capacitor  305 , as visible from EQN. 5, low sense node capacitance remains a very efficient means to reduce the equivalent input charge corresponding to other noise sources as e.g. the self-generated noise of the active buffer or the noise filtering circuit itself. 
         [0060]    A correctly dimensioned charge pulse detecting circuit according to an embodiment of the invention provides noise reduction factors ranging, for example, from greater than two to a hundred. The upper mentioned value of the noise reduction factor is defined by practical limitations explained in the following paragraphs. Typical values of RMS input charge variation equivalent to the output noise contributed by the recharge resistor range from, for example, two to a hundred holes or electrons. 
         [0061]    For a given high-pass frequency, which is substantially defined by the width of the pulses to be detected, and a minimized capacitance C s  of sense capacitor  305 , a practical limit to the noise reduction effect is usually found due to high required resistance values of recharge resistor  304 . 
         [0062]    Furthermore, reducing the recharge transition frequency, i.e. increasing duration of the recharge transition, decreases the maximum average repetition rate of the pulses that can be reliably detected, which may also impose an ultimate practical limitation to the possible noise reduction effect. 
         [0063]    Besides reducing the effect of the most dominant noise source of known charge pulse detecting circuits, the charge pulse detecting circuit according to the present invention also allows controlling the contributions of its remaining noise sources by proper dimensioning of its elements. 
         [0064]    Thermal noise of source follower transistor  302  and current source transistor  303  may be decreased by increasing their transconductance values, while keeping band-limiter transition frequency  404  constant. 
         [0065]    Thermal noise of high pass filter resistor  307  may be reduced e.g. by decreasing its resistance value and simultaneously increasing the capacitance of high-pass filter capacitor  306  by inverse proportion, in such a way that high-pass filter transition frequency  403  remains constant. 
         [0066]    Thermal noise from band-limiting source follower transistor  308  and current source transistor  309  may be reduced by increasing the capacitance value of band-limiting capacitor  310  as well as the transconductance of band-limiting source follower transistor  308  proportionally and keeping the transconductance of current source transistor  309  lower (e.g., by a factor of 0.5) than the transconductance of band-limiting source follower transistor  308 . 
         [0067]      FIG. 6  schematically depicts another embodiment of the present invention, where active buffer  202  provides voltage amplification and is implemented as a common source amplifier transistor  602  and an active load transistor  603 . Note that this implementation of active buffer  202  is an inverting amplifier, i.e. its AC output voltage has inverse polarity with respect to its AC input voltage. Active load transistor  603  is connected in a diode configuration and its W/L ratio is chosen to be smaller than the W/L ratio of common source amplifier transistor  602 , in order to provide open loop gain larger than unity. Note that the described implementation of active buffer  202  providing voltage amplification is not to be construed as limiting. Further examples of possible implementations include but are not limited to common source amplifiers with active current source loads, cascoded common source amplifiers with either diode connected or current source active loads and CMOS inverters. Note that different transistor types than schematically depicted in  FIG. 6  may be used to achieve equivalent functionality. 
         [0068]    In the described embodiment, signal charge delivered from a sensing device  601  onto a sense node  611  is removed across a recharge resistor connected between said sense node  611  and an amplifier output node  612 . The DC potentials on sense node  611  and amplifier output node  612  are thus set to essentially the same non-saturated operating voltages by feedback operation of recharge resistor  605  and the active buffer providing voltage amplification. 
         [0069]    In this embodiment the implementation of noise filter  206  may be essentially identical to the implementation depicted in  FIG. 3 . 
         [0070]    In the case of the discussed embodiment for a given capacitance Cs of the sense capacitor  605  and a given resistance R r  of a recharge resistor  604 , the recharge transition frequency is found to be Av/2πR r C s  where Av is the DC voltage amplification of the amplifier comprising common source amplifier transistor  602  and active load transistor  603 . Note that, as an effect of feedback operation, the recharge transition frequency is increased by a factor of the voltage amplification with respect to a recharge resistor connected to a DC potential rather than amplifier output  612 . 
         [0071]    For a given width of the charge pulses to be detected and a resulting maximum high-pass filter frequency, higher resistance values of recharge resistor  604 , than for instance in the embodiment of  FIG. 3 , may be needed in order to attenuate the noise due to recharge resistor  604  to a desired level. 
         [0072]    An important advantage of embodiments of charge detecting circuits employing an implementation of active buffer  202  providing voltage amplification larger than unity is, that excellent conversion factors of output voltage on amplifier output node  612  over input charge on sense node  611 , for example ranging from 50 micro-volts to 5 milli-volts per electron or hole, may be obtained. This reduces the input charge noise equivalent to the self-generated noise of the noise filter as well as further downstream readout circuitry  630 . Therefore, requirements for the self-generated noise of the noise filter circuit may be less stringent, while not compromising the overall noise performance of the detecting circuit. In particular, less semiconductor area may be needed for high-pass filter capacitor  606  and band-limiting capacitor  610 . Furthermore, the transconductance and thus the current consumption of band limiting source follower transistor  608  in the currently discussed embodiment might be reduced, for example by a factor ranging from two to the voltage gain of active buffer  202 , compared to band limiting source follower transistor  308  in the embodiment wherein active buffer  202  does not provide voltage amplification. 
         [0073]    Referring now to  FIG. 7 , an embodiment of the present invention is schematically depicted, wherein noise filter  206  is an active filter, and wherein noise filter  206  provides voltage amplification larger than unity to the signal edge, i.e. it provides voltage amplification in its passing band of frequencies. Recharge device  204  and active buffer  202  of the disclosed embodiment may be essentially identical to the corresponding elements of the embodiment schematically shown in  FIG. 3 . 
         [0074]    A noise filter  720  of this embodiment may comprise a high pass filter capacitor  706  with its terminals connected to a buffer output node  712  and an intermediate node  714 , a band-pass filter amplifier transistor  708  and an active load transistor  709  forming an inverting voltage amplifier amplifying the voltage on intermediate node  714  to an output node  713 , a band-limiting capacitor  710  connected to output node  713  and a high-pass filter resistor  707  connected between intermediate node  714  and output node  713 . 
         [0075]    A stable DC operating voltage on intermediate node  714  is established by feedback operation of the inverting voltage amplifier and high-pass filter resistor  707 . Active load transistor  709  is connected in a diode configuration and its W/L ratio is chosen to be smaller than the W/L ratio of band-pass filter amplifier transistor  708  in order to provide open loop gain greater than unity. Note once more that the implementation of this amplifier is not to be construed as limiting. Other amplifier types may be used without deviating from the scope of the present invention. Furthermore different transistor types than depicted in  FIG. 7  may be used to achieve equivalent functionality. 
         [0076]    The transfer function of noise filter  720 , i.e. the ratio of the voltage on output node  713  over the voltage on intermediate node  714  H nf (s) can be approximated by the expression (EQN. 6) below, under the assumption of sufficient transconductance gm 2  of band-pass amplifier transistor  708 , a resistance R hp  of high-pass filter resistor  707  larger than the DC output impedance Ro of the inverting voltage amplifier and neglecting the parasitic capacitance on intermediate node  714 . 
         [0000]    
       
         
           
             
               
                 
                   
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                         sgm 
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                         R 
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                             s 
                              
                             
                                 
                             
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                               C 
                               hp 
                             
                           
                         
                         ) 
                       
                        
                       
                         ( 
                         
                           1 
                           + 
                           
                             s 
                              
                             
                                 
                             
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                         ) 
                       
                     
                   
                 
               
               
                 
                   ( 
                   6 
                   ) 
                 
               
             
           
         
       
     
         [0077]    In EQN 6 R hp  is the resistance of high-pass filter resistor  707 , C hp  is the capacitance of high-pass filter capacitor  706  and C l  is the capacitance of band-limiting capacitor  710 . The above expression furthermore is based on the assumption that the product of R hp C hp  is larger than R o C l . 
         [0078]    We thus observe that noise filter  720  is a band-pass filter with a lower band limit a the frequency of ½πR hp C hp , the upper band-limit at the frequency of ½πR o C l  and a voltage amplification in the passing band of gm 2 R o , i.e. the voltage amplifier open loop gain. 
         [0079]    Therefore, for a given width of the charge pulses to be detected and a given desired attenuation of the thermal noise from recharge resistor  704 , the same dimensions of high-pass filter resistor  707  and high-pass capacitor  706  as in a noise filter without voltage amplification such as depicted in  FIG. 3  and  FIG. 6  may be used. 
         [0080]    If a MOS transistor is employed in strong inversion triode region and/or even weak inversion mode is used in order to implement high-pass filter resistor  707 , attention has to be paid to the proper biasing of the gate of said MOS transistor, in order to avoid excessive resistance variations. Due to the fact that the required resistance values of high-pass filter resistance are generally lower than, for instance, the required resistance of recharge resistor  609 , the implementation of the embodiment of  FIG. 7  may be less complex, in practice, than the implementation of the embodiment of  FIG. 6 , and high-pass filter resistor  707  may be subjected to lower relative resistance variation than, for instance, recharge resistor  609 . 
         [0081]    The embodiment of  FIG. 7  provides high conversion factor values of input charge on sense node  711  to output voltage on output node  713 , for example ranging from 50 micro-volts to 5 milli-volts per electron or hole, thanks to voltage amplification in noise filter  720 . Therefore, this embodiment offers excellent immunity against noise from downstream readout circuitry  730 . However, due to unity gain in the active buffer, as opposed to the embodiment shown in  FIG. 6 , attention has to be paid to noise from the noise filter  720  itself, just as in the embodiment schematically shown in  FIG. 3 . 
         [0082]      FIG. 8  schematically depicts yet another embodiment of the present invention, where the signal charge is removed from a sense node  811  across a low impedance path in regular or irregular time intervals, instead of slowly and continuously removing signal charge across a high impedance path after the arrival of every charge pulse the from a detector  801 . 
         [0083]    The noise filter, as well as the active buffer formed by common source a amplifier transistor  802  and active load transistor  803 , may be essentially identical to their respective counterparts in the embodiment of  FIG. 6 . The recharge device used in the present embodiment, however, is a reset switch transistor  804  with one of its drain/source terminals connected to sense node  811  and one drain/source terminal connected to the amplifier output node  812 . 
         [0084]    The gate terminal of reset switch transistor  804  is pulsed such that said reset switch transistor  804  is closed for a relatively short time, for example a duration in the range of the width of the detected pulses, in regular or irregular intervals, and it is left open in the periods in-between that may have a duration corresponding, for example, to 10 to 1000 times the width of the detected pulses. 
         [0085]    When reset switch transistor  804  is closed, signal charge is removed across the low impedance path provided by said reset switch transistor  804 , and non-saturating amplifier input and output voltages are established on sense node  811  as well as on amplifier output node  812  by negative feedback operation of the inverting amplifier and reset switch transistor  804 . Other transistor types than the depicted n-channel MOS transistor depicted in  FIG. 8  may be used to implement reset switch transistor  804 . 
         [0086]    Note that, if any signal charge is available on sense node  811 , a voltage pulse triggered by the closing of reset switch transistor  804  appears on an output node  813 . This output voltage pulse corresponding to the reset action has an amplitude depending on the amount of signal charge integrated on the sense node capacitor  805  between the previous reset action and the current reset action and a decay time corresponding to the product of the resistance of high-pass resistor  814  times the capacitance of the high-pass filter capacitor  806 . Note that the output voltage pulse corresponding to the reset action has a polarity that is opposite to the polarity of output voltage pulses caused by signal charge pulses from sensing device  801 . For example, if signal charge pulses comprise electrons, signal voltage pulses on output node  813  have positive polarity whereas reset voltage pulses on output node  813  have negative polarity. Thanks to their opposite polarities, reset pulses can be easily distinguished from signal pulses, besides the fact that the arrival time of reset pulses is well known. 
         [0087]    Also note that a dead time, during which no signal pulses can be detected, exists while reset switch transistor  804  is closed. 
         [0088]    The noise from reset switch transistor  804  observed at sense node  811  is frozen once said reset switch transistor  804  is opened, i.e. it mainly comprises a DC component and no substantial higher frequency spectral component. Such noise is, theoretically, completely eliminated by the high pass function of continuous time noise filter  206 . I should be noted that the term “completely” also encompasses the term “substantially completely”. 
         [0089]    Thanks to the efficient suppression of recharge noise and the excellent immunity against noise from downstream circuitry  830  and noise filter noise due to voltage amplification provided by the active buffer, this embodiment of the present invention offers overall RMS input charge variations equivalent to the overall output noise lower than, for example, ten electrons or holes.