Abstract:
An electric motor ( 10 ) which includes an armature ( 11 ) with at least two armature phase pair windings ( 12 ) and salient pole rotor arrangement ( 15 ) having field windings ( 17 ) terminating in a selective electrical switch which determines the electrical continuity of said field windings ( 17 ). Also included is control means which is configured to regulate the magnetizing of the field winding ( 17 ) so that, at any given moment, one armature phase pair is usable for magnetizing the field winding whilst the other pair is responsible for torque production.

Description:
FIELD OF THE INVENTION 
       [0001]    This invention relates to an electric motor. 
       BACKGROUND TO THE INVENTION 
       [0002]    A synchronous motor has inherent advantages over induction and DC motors. In a synchronous motor, the speed of the rotor is exactly proportional to the frequency of the system which supplies the synchronous motor with electrical power. Accordingly, the synchronous motor is further characterized by the fact that it runs at a constant speed (for constant supply frequency) at leading power factor and has low starting current. The efficiency of such a motor is generally higher than for other electric motor types. 
         [0003]    An armature of a synchronous motor is generally built with one set of AC poly-phase distributed windings, usually on the stator or outer annular ring of the motor. As such, the field winding of the motor is usually found on the rotor, and typically consists of more than one pole pair. It is generally understood that the field poles are excited with direct current. The configuration and method of exciting the rotor field windings determines the type of synchronous motor. In general, a separate exciter, slip rings, and brushes are required. 
         [0004]    The exciter, slip rings and brushes are eliminated in the synchronous induction motor which has a rotor designed with differing reluctance paths across the air gap separating the stator and rotor to facilitate in developing reluctance torque. There are no field windings on the rotor, and hence no excitation source is required. The stator armature windings are then powered directly from an AC supply line. 
         [0005]    The induction motor is simple and cheap to manufacture but does not offer the performance of the synchronous motor. On the other hand, slip-ring synchronous motors are more complex and expensive because of the exciter, slip rings and brushes required for operation. This also increases maintenance requirements, whereas the induction motor is virtually maintenance free. Brush DC motors have similar drawbacks although they are easier to control and have excellent traction characteristics. The slip rings and brushes are eliminated in brushless synchronous motors but the AC brushless exciter configuration still means additional cost, space and complexity. 
         [0006]    Permanent magnet synchronous motors appear to be the most attractive. However, the manufacture and high cost of high field strength permanent magnets, and the process of attaching these magnets to the rotor, especially for large machines, becomes an engineering challenge. The maximum field strength of permanent magnets is also limited by the current state of the art. The synchronous induction motor, although very simple in construction, is not very efficient and is generally much larger than a slip-ring synchronous motor for similar performance. In practice, synchronous induction motors have not found much use above a few kilowatts. 
       SUMMARY OF THE INVENTION 
       [0007]    According to an aspect of the invention there is provided an electric motor which includes 
         [0008]    an armature having at least two armature windings arranged to form two respective armature phases; 
         [0009]    a salient pole rotor arrangement within the armature which rotor is separated from the armature by an air gap, said rotor being shaped and configured so that the salient pole arrangement defines a higher and a lower magnetic reluctance path through said gap between the rotor and armature, and which rotor further includes field windings about the salient poles of the rotor which windings are configured to form at least one pole pair, said field windings terminating in a selective electrical switch which determines the electrical continuity of said field windings so that a reverse voltage bias imposed across the switch results in the field winding being open circuit; and 
         [0010]    control means which is configured to regulate the magnetizing of the field winding by applying a voltage to the armature phase linked to the field winding via the lower reluctance path so that the energized armature phase in turn imposes a reverse voltage bias across the switch through induction, this reverse bias preventing the flow of current in the field winding but the applied voltage to the armature phase increasing magnetic flux density in the rotor, the control means thereafter removing said applied voltage once a predetermined flux density in the rotor is reached so that the removal of applied voltage induces a reversal of voltage to a forward voltage bias across the switch allowing current to flow in the field winding which current prevents the decay of the flux density in said rotor, and which control means is further configured to regulate the production of torque in the motor by applying a voltage to the other respective armature phase not responsible for energizing the field winding, so that, at any given moment, one armature phase is usable for magnetizing the field winding whilst the other phase is responsible for torque production. 
         [0011]    It is to be appreciated that the selective switch in the field winding improves the efficiency of the motor by effectively capturing the magnetic flux density in the rotor when current is allowed to flow in the field winding. This no longer requires the armature winding to supply magnetizing current continuously as well as torque current, as is the case with, for example, existing induction motors. 
         [0012]    It is further to be appreciated that the motor employs a switched mode flyback principle in order to magnetize the rotor. 
         [0013]    The armature may include a plurality of packed slotted metal laminations to reduce eddy-currents in the armature. The rotor may include a plurality of packed slotted metal laminations to reduce eddy-currents in the rotor. 
         [0014]    The selective switch may include a freewheeling diode. The selective switch may include a solid-state device, e.g. a transistor, a thyristor, or the like. 
         [0015]    The control means may include a microprocessor. The control means may include electronic switches for controlling the energizing of the armature phases. The electronic switches may include transistors. Accordingly, the electronic switches may be arranged in an H-bridge arrangement. 
         [0016]    The control means may include sensors for sensing the position of the rotor relative to the armature phases so as to regulate the energizing of the phases at the correct instances. 
         [0017]    The control means may be configured to determine the position of the rotor from armature phase current and voltage characteristics. 
         [0018]    The control means may regulate the flux density in the rotor depending on the speed of the motor, e.g. at high speed the magnetization of the field winding need only be topped up every few revolutions of the rotor, whereas at low speed the field winding may require topping up multiple times during one revolution of the rotor. 
         [0019]    The control means may be configured to control the motor as a generator under suitable circumstances. 
         [0020]    An armature phase may be dedicated to magnetizing the field winding. Accordingly, an armature phase may be dedicated to torque production in the motor. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0021]    The invention is now described, by way of non-limiting example, with reference to the accompanying drawings wherein 
           [0022]      FIG. 1  shows, in radial cross-sectional view, an electric motor, in accordance with the invention; 
           [0023]      FIG. 2  shows, in axial cross-sectional view, the electric motor of  FIG. 1 ; 
           [0024]      FIG. 3  shows, in schematic view, a circuit diagram representing an armature phase interaction with the field winding through the lower magnetic reluctance path; 
           [0025]      FIGS. 4   a  and  4   b  show, in radial cross-sectional view, the rotor of the motor in different positions relative to the armature; 
           [0026]      FIG. 5  shows graphs of the current and flux density of the different field and armature windings during operation of the motor shown in  FIG. 1 ; 
           [0027]      FIG. 6  shows, in schematic view, one embodiment of the control means; 
           [0028]      FIG. 7  shows, in radial cross-sectional view, a further embodiment of the motor; and 
           [0029]      FIG. 8  shows graphs of the armature winding currents during operation of the motor shown in  FIG. 7 . 
       
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       [0030]    With reference to the accompanying drawings, an electric motor, in accordance with the invention, is generally indicated by reference numeral  10 . 
         [0031]    Referring to  FIGS. 1 and 2 , the outer member of the motor  10  consists of packed slotted steel laminations  11  for holding the armature windings  12 , which together constitute the stator  13 . The armature windings  12  are further divided into two separate windings  12 . 1  and  12 . 2  constituting two separate phases. 
         [0032]    Similarly, the inner member consists of packed steel laminations  15  relatively fixed to a drive shaft  16  with a field winding  17  terminating in the selective switch being a freewheeling diode or rectifier  18  in this embodiment which collectively constitute the rotor  14  being able to rotate relative to the stator  13 . 
         [0033]    It is to be appreciated that, for the purposes of explanation, a reference coordinate system d-q is shown on  FIG. 1 . This coordinate system is rotatively associated or referenced to the rotor  14 . Regarding the salient pole arrangement of the rotor  14 , the air gap in the direct axis d is much smaller than the air gap in the quadrature axis q. 
         [0034]    The reluctance of the magnetic circuit along the q axis is thus much greater than the reluctance along the d axis. This is an important feature of the invention and is integral to its mode of operation. The higher reluctance in the q axis reduces the magnetic coupling of the torque producing armature phase with the rotor  14 , which is desirable. The larger air gap along the q axis also provides space for the field winding  17  resulting in the salient pole structure. 
         [0035]    It is to be appreciated that the invention generally requires the rotor  14  to be of a salient pole nature. The shape of the laminations  15  are similar to that of a reluctance motor (not shown), but the motor  10  does not operate on reluctance torque; the action of the field winding  17  and diode  18  combination are directly involved in the production of torque. 
         [0036]    In the embodiment shown, the rotor  14  has one pole pair, but in other embodiments, the rotor may be constructed with multiple pole pairs. Similarly, the stator  13  may be constructed with more than two phases in further embodiments. 
         [0037]    Referring now to the position of the rotor  14  relative to the armature phase windings  12 . 1  and  12 . 2  in  FIG. 1 . One novel aspect of the invention is the method by which the rotor  14  pole pair is magnetized, i.e. the method by which the field winding  17  is excited. Ignoring the action of the field winding  17  for the moment; current applied to the armature phase  12 . 1  will produce a magnetic field in the direction of the q axis. Similarly, current applied to the armature phase  12 . 2  will produce a magnetic field in the direction of the d axis. The magnetic field in the d axis will be many times larger than that in the q axis due to the much larger reluctance in the q axis. 
         [0038]    In operation, either stationary or rotating, by correct timing and switching of the armature phases  12 . 1  and  12 . 2  the magnetic field excitation of the rotor  14  will be kept at a maximum along the d axis and a minimum along the q axis (essentially zero). 
         [0039]    When the armature phase winding  12 . 1  or  12 . 2  centre axis is aligned with the rotor  14  direct axis d, then the particular armature winding is strongly coupled to the field winding  17 . Similarly, when the armature phase winding  12 . 1  or  12 . 2  centre axis is aligned with the rotor  14  quadrature axis q, then the particular armature winding is weakly coupled to the field winding  17 . 
         [0040]      FIG. 3  shows a schematic of a transformer representation  20  of the armature winding  12 . 2  strongly coupled to the field winding  17  terminated in the diode  18 . The armature winding  12 . 2  serves as the primary of the transformer  20 , and the field winding  17  serves as the secondary of the transformer  20 . It is to be appreciated that the transformer representation  20  is only valid when the centre axis of the armature phase winding  12 . 2  is aligned with the rotor direct axis d, i.e. strongly coupled. When the armature winding  12 . 2  is in quadrature to the field winding  17 , there is no transformer action due to the respective windings&#39; central axis being perpendicular to one another. 
         [0041]    As is convention, the black dots  21 . 1  and  21 . 2  indicate the “same” polarity of the respective windings. There are two positions of the rotor  14  in one full cycle of rotation when the armature phase  12 . 2  centre axis aligns with the rotor d axis. 
         [0042]    In a first position, with the switch  22  closed so that positive voltage is applied to the primary  12 . 2 , the diode  18  will be reverse biased as shown in  FIG. 3 . In a second position, the rotor  14  being 180° rotated with positive voltage applied to the primary  12 . 2 , the diode  18  will be forward biased (not shown). 
         [0043]    Considering the first position, the switch  22  is closed thus applying a positive voltage to the primary winding  12 . 2 . A voltage is induced in the secondary  17  which applies a reverse bias to the diode  18  preventing current from flowing in the secondary winding  17 . However, the magnetizing current in the primary winding  12 . 2  and, hence, the magnetic flux density in the rotor laminations  15  will increase or ramp up from zero, according to: 
         [0000]        v=L·di/dt   (1) 
         [0000]        di=dt·v/L   (2) 
         [0000]    where:
       v is the applied voltage in volts (V)   L is the primary inductance in Henrys (H)   di is the change in current in amps (A)   dt is the change in time in seconds (s)       
 
         [0048]    The switch  22  is opened when the magnetic flux density reaches the predetermined or desired value. This interrupts the primary current which must decrease to zero—the rate of change of the current di/dt is therefore negative and from equation (1) the voltage across the primary winding becomes negative. 
         [0049]    Accordingly, the secondary winding  17  also experiences a voltage reversal through induction, causing the diode  18  to become forward biased and conducting. The current thus caused to flow in the secondary winding  17  will be proportional to the magnetic flux density created in the core laminations  15  by the primary winding  12 . 2  in the first place. The current in the rotor field windings  17  effectively captures the magnetic flux density in the core laminations  15 . 
         [0050]    It is to be appreciated that the voltage impressed across the secondary winding  17  by the diode  18  forward voltage is much lower than the voltage initially applied across the primary  12 . 2 . From equation (1), this means that the decay rate of the secondary  17  current di/dt and hence the decay of the magnetic flux density is much slower than the ramp rate of the primary  12 . 2  current, and hence the ramp rate of the magnetic flux density. 
         [0051]    As such, the decay time of the field magnetization can be designed to be orders of magnitude longer than the ramp time by the correct choice of applied voltage, and primary to secondary turns ratio. For example, a typical applied voltage may be 300 V and diode forward voltage may be 1V, and accounting for the turns ratio, once the rotor field is at maximum magnetization it only needs to be charged for 50 us every 15 ms, say, to maintain the field strength within 10% of its maximum. 
         [0052]    Considering now then only the magnetization of the rotor  14  of  FIG. 1  (and not torque production) with the rotor  14  rotating at speed relative to the stator  13 . The control means or power-drive-electronics (not shown) briefly applies the correct polarity voltage to the particular armature phase  12 . 1  or  12 . 2  whenever the rotor  14  direct axis d (with the correct polarity of diode,  18 ) aligns momentarily with, in passing, the phase  12 . 1  or  12 . 2  centre axis to give the rotor  14  a magnetic “charge”. 
         [0053]    At start up, the magnetic “charge” time will be ten times longer, say, than the running top up “charge” time, in order to get the rotor  14  magnetic field up to its maximum value initially. When running at high speed the rotor  14  magnetization may only need topping up every few cycles whereas at very low speed it may need topping up a few times per cycle. 
         [0054]    In a preferred embodiment of the invention, a diode  18  is used as the switching element in the rotor winding  17 . It is however to be appreciated that any type of switching element that is suitably synchronized to the switching of the armature phase windings  12 . 1  and  12 . 2  may be used, e.g. transistor, thyristor, MOSFETs, and/or the like. 
         [0055]    A person skilled in the art will appreciate that a similar principle to that described above is employed in the operation of a flyback transformer in a switched mode power supply. However, in the flyback transformer, energy is continuously transferred from the primary winding to a fixed secondary winding connected to a resistive load, whereas in the motor  10  the flyback principle is used to maintain the magnetic flux density in a relative, rotating rotor. 
         [0056]    Let us now consider torque production in the motor  10 . The force exerted on a current carrying conductor in and perpendicular to a magnetic field is given by the Lorentz force equation: 
         [0000]        F=I·i·B   (3) 
       Where: 
       [0000]    
       
         
           
             F is the force in Newtons, N. 
             I is the conductor length in metres, m. 
             i is the current in the conductor in amps, A. 
             B is the magnetic flux density in Tesla, T.
 
And the torque in a motor would be given by;
 
           
         
       
     
         [0000]    
       
         
           
             
               
                 
                   
                     
                       
                         T 
                         = 
                           
                          
                         
                           F 
                           · 
                           r 
                         
                       
                     
                   
                   
                     
                       
                         = 
                           
                          
                         
                           l 
                           · 
                           i 
                           · 
                           B 
                           · 
                           r 
                         
                       
                     
                   
                 
               
               
                 
                   
                     
                       
                         ( 
                         
                           From 
                            
                           
                               
                           
                            
                           
                             ( 
                             3 
                             ) 
                           
                         
                         ) 
                       
                     
                   
                   
                     
                       
                         ( 
                         4 
                         ) 
                       
                     
                   
                 
               
             
           
         
       
     
       Where: 
       [0000]    
       
         
           
             r is the rotor radius. 
             T is the torque in N·m. 
             F is the force in Newtons, N. 
           
         
       
     
         [0064]    Referring to  FIG. 1 , assume that the rotor  14  is fully magnetized. The magnetic flux lines  19  passing azimuthally through the stator core  13  do not cross the conductors of the armature phase  12 . 2  and hence these conductors experience no force. The conductors of armature phase  12 . 1  however, are perpendicular to, and lie directly in, the magnetic flux lines  19  crossing from the stator  15  to the rotor  14  across the air gap and thus experience a torque according to equation (4). 
         [0065]    Since the conductors of the armature phase winding  12 . 1  are fixed in the stator  13  and since the rotor  14  in turn experiences an opposite reaction, the rotor  14  will experience the resultant torque and motion. Thus torque is produced in the rotor  14  by current flowing in the stator conductors  12 . 1  or  12 . 2  that are located in the smaller, or d axis, air gap. 
         [0066]    In operation, both the armature phases  12 . 1  and  12 . 2  will alternately produce torque and magnetizing charge as the motor rotates. The torque producing current will be applied for a large portion of the rotor cycle whereas magnetization will be a fraction of the time. Microprocessor controlled power transistors, with suitable position detection of the rotor  14 , and armature phase current feedback, can accomplish the required timing and current control between the phases. 
         [0067]    The angular position of the rotor  14  can be determined directly through sensing elements mounted on the rotor  14  and feedback circuits. Otherwise, the position can be indirectly determined from the voltage and current characteristics of the armature windings since these are affected by the variable reluctance presented by the rotor  14  dependent on its angular position (due to the difference in air gap in the direct and quadrature axis). 
         [0068]    In other words, in a first rotor position shown in  FIG. 4   a , armature phase winding  12 . 1  produces torque when carrying current while at the same time armature phase winding  12 . 2  charges up the magnetic field in the rotor  14 . Similarly, in a second rotor position shown in  FIG. 4   b , the roles are reversed and armature phase winding  12 . 2  produces torque while armature phase winding  12 . 1  charges up the magnetic field in the rotor  14 . 
         [0069]    Graphs showing the currents I 12.1 , I 12.2 , and I 17  for the armature phase windings  12 . 1 ,  12 . 2 , and the rotor field winding  17 , respectively, and the rotor  14  magnetic flux density B 14  verses time over one full cycle of rotation are given in  FIG. 5 , for a particular direction of rotation. The armature phase current I 12.1  and I 12.2  waveforms consist of two torque producing segments and two magnetizing pulses in a full cycle, as indicated. 
         [0070]    It is to be appreciated that the direction of rotation of the rotor  14  is easily reversed by changing the polarity of one of the armature phase currents, I 12.1  or I 12.2 , or simply by swapping the timing sequence of I 12.1  and I 12.2 . 
         [0071]    The motor  10  will operate as a generator by reversing the polarity of the armature phase current I 12.1  and I 12.2  torque producing segments, but with the same timing and polarity of the magnetizing current pulse as they are shown in  FIG. 5 . 
         [0072]    Whether motoring or generating, in forward or reverse direction, the timing of the currents are synchronized to the rotor  14  position. 
         [0073]    An electronic circuit for the control means for driving the motor  10  is shown in  FIG. 6 . A schematic representation of the motor  10  is included showing the armature windings  12 . 1  and  12 . 2 ; the rotor  14 , the rotor field winding  17 , and the free wheeling diode  18 . 
         [0074]    In the embodiment shown, a DC power source (not shown) supplies the main DC bus  45 . 1  and  45 . 2  for the motor  10 . An “H-bridge” arrangement of MOSFET transistors  40 . 1 ,  40 . 2 ,  40 . 3 ,  40 . 4  is used to switch the required current to the armature phase winding  12 . 1  and an “H-bridge” arrangement of MOSFET transistors  42 . 1 ,  42 . 2 ,  42 . 3 ,  42 . 4  is used to switch the required current to the armature phase winding  12 . 2 . 
         [0075]    This allows either phase to be independently controlled, for positive and negative current polarity and, through the employment of pulse width modulation, independent control of the current magnitudes. 
         [0076]    A microprocessor  52  controls the switching of the MOSFETs via a MOSFET driver interface  50 , and hence directly controls the timing, magnitude, and polarity of the currents I 12.1 , I 12.2  and I 17 . Current sensor elements  41 . 1 ,  41 . 2 ,  41 . 3 ,  41 . 4  provide feedback signals to the microprocessor  52 , via an analog to digital converter  54 , for current magnitude control. 
         [0077]    The position of the rotor  14  relative to the stator  13  is required for the microprocessor to control the phase currents, I 12.1  and I 12.2  and rotor field current I 17 , in magnitude and timing, according to  FIG. 5 . An example of a rotor position detecting means is shown by use of a disc  30  rotatively fixed to the rotor  14 ; and four optically reflective sensors  31 . 1 ,  31 . 2 ,  31 . 3 ,  31 . 4  fixed relative to the stator of the motor. The disc  30  has a 90° segment reflectively coloured (white) as shown, with the remaining 270° segment being non-reflective. 
         [0078]    The signals from the optical sensors  31 . 1 ,  31 . 2 ,  31 . 3 ,  31 . 4  are returned to the microprocessor  52  via a buffer, or signal conditioning element  56 . With the disc  30  positioned as shown with the white, reflective element under optical sensor  31 . 2 , the signal returned by optical sensor  31 . 2  would be a digital “1” for example, and the signals returned by the remaining sensors  31 . 1 ,  31 . 3 ,  31 . 4  would be a digital “0”. The microprocessor  52  thus controls the current applied to the armature phase windings  12 . 1  and  12 . 2  in accordance with  FIG. 5  synchronized to the rotor  14  position, determined by the digital combinations returned by the optical sensors  31 . 
         [0079]    Other detection means may also be employed, such as magnetically coupled position detection, direct slide contact switches, commutator, or the like. 
         [0080]    In a preferred embodiment, the microprocessor  52  monitors the state of switches and a variable resistor included in the user interface panel  58  to start, stop, or change direction of the motor. Upon starting the motor  10 , the rotor  14  is initially completely unmagnetized, i.e. no current I 17  flows through the field winding  17  and free wheeling diode  18 . 
         [0081]    As such, the rotor  14  must first be fully magnetized or “charged” by whichever is strongly coupled of the armature windings  12 . 1  or  12 . 2  to the rotor field winding  17  (as per  FIG. 3  and its corresponding description). The microprocessor  52  determines the orientation of the rotor  14  with respect to the armature windings  12 . 1  and  12 . 2  from the signals obtained from sensors  31  and selects the winding  12 . 1  or  12 . 2  which is strongly coupled to the rotor field  17  and determines the required current direction. The microprocessor  52  switches on the required MOSFET transistor pair from the two “H-bridges”  40  and  42  to charge up the magnetic flux density B 14  in the rotor  14  to the required value. 
         [0082]    The rotor  14  remains stationary during this process. Once the magnetic flux density has reached the required value, the active or charging armature winding of  12 . 1  or  12 . 2  is switched off. The previously idle armature winding of  12 . 1  or  12 . 2  is then switched on to produce torque. If there is a reasonable or light load coupled to the motor  10  it will accelerate up to speed and the timing and control of currents will resemble those shown in  FIG. 5 . However if the load is too large for the motor  10 , i.e. locked rotor  14 , or a high inertia load, the first armature winding of  12 . 1  or  12 . 2  will have to keep topping up the rotor  14  magnetic flux density B 14 , while the second winding will have to keep supplying driving or torque current continuously until the rotor  14  begins turning whereupon the two armature windings  12 . 1  and  12 . 2  will begin alternately supplying torque current and magnetizing current according to the rotor  14  position. 
         [0083]    The above general description and mechanicals, windings and electronics are the preferred embodiment of the invention. A second embodiment could be implemented with armature windings  12 . 1  and  12 . 2  being always torque-producing windings, with additional windings  12 . 3  and  12 . 4  located in the stator being always magnetizing windings. A schematic of this embodiment is shown in  FIG. 7 , and the corresponding current graphs are shown in  FIG. 8 . 
         [0084]    The graphs in  FIG. 5  are an example of possible armature currents I 12.1  and I 12.2  wave-shapes and timing. The torque producing current portions are shown as constant values for a quarter of a cycle, or  900  of mechanical angle of the rotor  14 , for simplicity. These wave-shapes may be modified to optimize torque and reduce harmonics e.g. stepped, sinusoidal, or the like. 
         [0085]    The magnetizing current pulse portion of the armature currents I 12.1  and I 12.2  may have a much larger magnitude than the torque producing portion but the pulse width will be a hundred times, or more, narrower resulting in an average or RMS value of magnetizing current far less than the average or RMS value of torque current. 
         [0086]    For example, the magnetizing pulse RMS value may be only 0.5 A for an RMS torque current value of 10 A. The magnetizing pulse current therefore produces very little heating losses in the stator windings  12  compared to the torque producing current. The major portion of magnetizing losses occurs in the rotor windings  17 , since the rotor field winding current I 17  is very nearly continuous at a reasonable value. 
         [0087]    It is to be appreciated that the maximum magnetization current in an induction motor is generally limited by the current carrying capacity of the stator windings and harmonic distortion of the sinusoidal magnetizing current when approaching magnetic saturation. The maximum magnetization of a permanent magnet motor is limited by the current state of the art concerning magnet production. 
         [0088]    The motor  10  herein described does not suffer from these drawbacks and a much higher magnetization of the rotor and higher torque current is obtained, at comparable efficiencies, resulting in higher power density and torque for similar sized motors. 
         [0089]    The overall efficiency of the motor  10  can be dynamically optimized by keeping the rotor  14 &#39;s copper losses equal to the stator  13 &#39;s copper losses. The microprocessor  52  can adjust the magnitudes of torque current and magnetizing current for optimum efficiency, over the full speed range and variable loading. 
         [0090]    It shall be understood that the examples are provided for illustrating the invention further and to assist a person skilled in the art with understanding the invention and is not meant to be construed as unduly limiting the reasonable scope of the invention. 
         [0091]    The Inventor regards it as an advantage that the motor has overall performance exceeding that of the synchronous motor, induction motor and brush or brushless DC motor, whilst incorporating the simplicity and cost effective manufacturability of the induction motor. 
         [0092]    The Inventor regards it as a further advantage that the motor has a higher power density and efficiency compared with all other motor types. The Inventor regards it as a yet further advantage that the motor has high starting torque for low starting current which finds particular application in traction applications. 
         [0093]    The Inventor also regards it as an advantage that dynamic speed and torque control at optimal efficiency over the full load range, with full four-quadrant operation (motoring, generating, forward and reverse), is simple to implement.