Abstract:
An improved method of commutating an electronically commutated motor ( 20 ) is disclosed. The motor has a positive ( 34 ) and a negative connecting lead ( 36 ), a stator having at least one winding phase ( 22 ), a permanent-magnet rotor ( 26 ), a rotor-position sensor ( 28 ) for producing a rotor position signal (u 1 , u 2 ), and a full bridge circuit ( 78 ), comprising a pair of Metal Oxide Semiconductor Field Effect Transistors (MOSFETs) coupled to each end of each winding phase ( 22 ). In order to prevent current shoot-through and prolong the service life of the MOSFETs, without employing a microprocessor, a respective-comparator ( 90, 90′, 92, 92 ′) is coupled to the gate of each MOSFET, in order to assure that the MOSFET is switched ON and OFF at an optimum time.

Description:
This application is a § 371 of PCT/EP03/00834, filed 28 Jan. 2003. 
   FIELD OF THE INVENTION 
   The invention concerns a method for commutating an electronically commutated motor, and a motor for carrying out such a method. 
   BACKGROUND 
   Electronically commutated motors operate with a stator winding that has a small number of phases. The motors principally used have one, two, or three phases. These phases can have power supplied to them in different ways depending on the type of electronics; supplying power via a full bridge circuit offers particular advantages. 
   With a full bridge circuit, steps must be taken to ensure that a short circuit does not occur in the full bridge. There are numerous circuits for this, for example the circuit according to U.S. Pat. No. 4,376,261, von der Heide, et al., which is based on the principle of creating a small gap between two control pulses. When the full bridge is controlled via a microprocessor (μP), corresponding current off-times can thus be “built into” the program, ensuring that upon commutation the one transistor switches off, then there is a delay of, for example, 50 μs, and only then is the other transistor switched on. A prerequisite for this, however, is that a microprocessor be used, and this is too expensive for many applications. The current off-time of, for example, 50 μs must furthermore be made so long that is sufficiently long under all operating conditions, with the result that, especially at higher rotation speeds, power output may be “wasted” because the current off-time could be shorter under many operating conditions. 
   SUMMARY OF THE INVENTION 
   It is therefore an object of the invention to make available a novel method for commutating an electronically commutated motor, and a motor having a full bridge circuit for carrying out such a method. 
   According to the invention, this object is achieved by feeding a rotor position signal to a comparator which controls when a bridge transistor turns OFF, monitoring the gate potential of the transistor being turned off, to determine when to turn ON the next transistor, and using another comparator to turn ON that next transistor. 
   A comparator is thus associated with each MOSFET of the bridge. When one of these MOSFETs is to be switched off, this occurs directly by way of its comparator. When one of these MOSFETs is to be switched on, the associated comparator then monitors the gate potential of the MOSFET that is to be switched off, i.e., in practice, its source-gate voltage. A reliable signal is thereby obtained as to when that MOSFET is blocked, and only then is the MOSFET that is to be switched on, switched on. It is very advantageous that with this method, the comparators are utilized differently depending on their instantaneous function (switching off or switching on), i.e. in one case as an amplifier, in the other as a comparator. The power output of such a motor can moreover be increased in many cases by optimizing the current off-time. 
   The stated object is achieved in a different fashion by controlling current flow through the winding phase using a p-channel MOSFET between the positive lead and the winding, and an n-channel MOSFET between the winding and the negative lead, with each MOSFET being controlled by a respective comparator, one of the comparators responding to the rotor position signal, and the other of the comparators responding to a gate voltage of the transistor being switched off. A motor of this kind combines reliable commutation with a very simple and inexpensive configuration, good power output, quiet operation, and a long service life for the MOSFETs in the full bridge circuit. 
   Further details and advantageous refinements of the invention are evident from the exemplary embodiment, in no way to be understood as a limitation of the invention, that is described below and depicted in the drawings. 

   
     BRIEF FIGURE DESCRIPTION 
       FIG. 1  is a circuit diagram of a preferred embodiment of an electronically commutated motor for carrying out a method according to the present invention; 
       FIG. 2  is a diagram showing pulse sequences u 1  and u 2 , to explain  FIG. 1 ; and 
       FIG. 3  is a circuit diagram to explain, by way of example, the generation of two opposite-phase rotor position signals u 1 , u 2 , the amplitudes of which depend on the operating voltage U B  of the motor. 
   

   DETAILED DESCRIPTION 
     FIG. 1  shows, for explanation of the invention, a single-phase two-pulse electronically commutated motor  20  having a single stator winding phase  22 , also having a permanent-magnet rotor  26  that is depicted as a four-pole rotor, and having a Hall generator  28  whose output signal is fed to an arrangement  30  having an output  32  and an output  33  antivalent thereto. In arrangement  30 , the output signal of Hall generator  28  is converted into two square-wave signals u 1  and u 2 , which are depicted in  FIG. 2  and proceed in oppositely-phased fashion. These can, if necessary, be shifted in phase in known fashion as a function of the rotation speed of motor  20 ; this is not depicted. 
   Motor  20  is supplied with voltage via a positive connecting lead  34  and a negative connecting lead  36  that is usually connected to ground  38 . Leads  34 ,  36  can be connected via respective connecting terminals  35  and  37  to a battery  40  whose voltage is labeled U B  and is usually, in this case, in the range between 5 and 32 V. The leads can also be powered from a rectifier, and are also referred to as a “DC link” circuit. 
   Located between leads  34  and  36  is a voltage divider having, for example, two identical resistors  44 ,  46 ,  50  that a potential of approximately U B /2, i.e. slightly less than half the operating voltage, is present at their connecting point  48 . The amplitude of pulses u 1  and u 2  is a function of U B  and is preferably also equal to U B /2. 
   Node  48  is connected to the anode of a diode  50  whose cathode is connected via a node  52  and a resistor  54  to negative lead  36 . A potential of approximately
 
( U   B /2−0.7 V)  (1),
 
is thus obtained at node  52 , i.e. lower than the potential U B /2 by an amount equal to the threshold voltage of diode  50  (approx. 0.7 V); this means that when operating voltage U B  changes (for example, because battery  40  is deeply discharged), the potential U B /2 at node  48  changes and the potential (U B /2−0.7 V) at node  52  likewise changes, so that these potentials change but a difference between nodes  48  and  52  of approx. 0.7 V is maintained.
 
   Node  48  is also connected via a resistor  58  and a node  60  to the anode of a diode  62  whose cathode is connected to negative lead  36 . The result is a potential at node  60  that is approximately 0.7 V higher than the potential (0 V) of lead  36 . 
   Four MOSFET transistors serve to control the current in phase  22  of motor  20 , namely a p-channel MOSFET  70  at upper left, a p-channel MOSFET  72  at upper right, an n-channel MOSFET  74  at lower left, and an n-channel MOSFET  76  at lower right. The four transistors  70  through  76  together constitute a full bridge circuit  78  in the form of an H-bridge. This is constituted by two half bridges, namely the left MOSFETs  70  and  74  on the one hand, and the right MOSFETs  72  and  76  on the other hand. Source S of transistors  70  and  72  is connected to positive lead  34 . Drain terminals D of transistors  70  and  74  are connected to a node  80  and to a terminal of winding phase  22 . Drain terminals D of transistors  72  and  76  are connected to a node  82  and to the other terminal of winding phase  22 . Source S of transistor  74  is connected via a resistor  84  to negative lead  36 , and source S of transistor  76  thereto via a resistor  86 . Resistors  84  and  86  can serve for current measurement, e.g. for a current limiter (not depicted). 
   If this is not desired, these resistors can be omitted. 
   When transistors  70  and  76  are conductive, a current flows from positive lead  34  through transistor  70  and terminal  80  to phase  22 , and on through transistor  76  and resistor  86  to negative lead  36 . When transistors  72  and  74  are conductive, on the other hand, a current then flows from positive lead  34  through transistor  72 , terminal  82 , winding phase  22 , transistor  74 , and resistor  84  to negative lead  36 . 
   During the switchover between these two states, it must not happen that transistors  70  and  74  are simultaneously conductive for a short period of time, or that transistors  72  and  76  are briefly both conductive, since the resulting short-circuit (or “shoot-through”) current would destroy these transistors or at least shorten their service life. The present invention is intended to prevent or at least greatly reduce this. 
   A comparator  90  serves to control upper left transistor  70 , and a comparator  92  to control lower left transistor  74 . In  FIG. 1 , the right half of the circuit is symmetrical to the left half, and the components there are therefore labeled identically but with an appended apostrophe (′). Upper right transistor  72  is thus controlled by a comparator  90 ′, and lower right transistor  76  by a comparator  92 ′. (The additional components on the right side are not described. For them, the reader is referred to the description of the left side of  FIG. 1 .) 
   Gate G of transistor  70  is connected via a capacitor  96 , and gate G of transistor  72  via a capacitor  96 ′, to positive lead  34 . Gate G of transistor  74  is connected via a capacitor  98 , and gate G of transistor  76  via a capacitor  98 ′, to negative lead  36 . These capacitors prevent abrupt changes in the voltage between source (S) and gate (G) of transistors  70  through  76  and, depending on their size, cause switching operations to become slower with the result that motor  20  runs more quietly. 
   Switching P-Channel Transistors  70  and  72  on and Off 
   When one of transistors  70  or  72  is to be switched on, its gate potential must be modified in the direction toward negative lead  36 ; in other words, its source-gate voltage USG must be increased to a value in the preferred range 1.5–4 V, maximum 20 V. In this case output  100  of upper comparator  90 , or output  100 ′ or upper comparator  90 ′, is therefore connected internally to negative lead  36 , thus causing voltage USG to increase correspondingly because the potential at output  100  substantially determines the potential at the gate of transistor  70 . 
   Conversely, when one of transistors  70  or  72  is to be switched off, its gate potential must be modified in the direction toward positive lead  34 , so that USG becomes less than 1.4 V. 
   This is done by making output  100  or  100 ′ high-resistance. 
   The potential at gate G of the relevant transistor  70  or  72  thus changes in the positive direction during the transition from the switched-on to the switched-off state, and when USG falls below a predetermined value, this means that the relevant transistor  70  or  72  is safely blocked, i.e. is in a high-resistance state. 
   Switching N-Channel Transistors  74  and  76  on and Off 
   When one of transistors  74  or  76  is to be switched on, its gate potential must be modified in the direction toward positive lead  34 , i.e. its gate-source voltage UGS must increase to a preferred value in the range 1.5 to 4 V, maximum 20 V. In this case, output  102  of lower comparator  92 , or output  102 ′ of lower comparator  92 ′, is therefore made high-resistance, with the result that voltage UGS of the relevant transistor  74  or  76  rises, and the latter becomes conductive. 
   Conversely, when one of transistors  74  or  76  is to be switched off, output  102  or  102 ′ is then connected to negative lead  36  so that UGS drops below 1.4 V and the relevant transistor  74  or  76  is blocked. 
   The potential at gate G of the relevant transistor  74  or  76  thus changes in the negative direction during the transition from the switched-on to the switched-off state; and when UGS falls below a predetermined value, this means that the relevant transistor  74  or  76  is blocked, i.e. is in its high-resistance state. 
   Negative input  104  of upper comparator  90 , like negative input  104 ′ of comparator  90 ′, is connected to node  60 , i.e. is at a potential of approximately 0.7 V with reference to negative lead  36 . Positive input  106  of comparator  90  is connected to the cathode of a diode  108 , and also via a resistor  110  to gate G of lower left transistor  74 . This gate G is in turn connected via a resistor  112  to output  102  of lower left comparator  92 , and via a resistor  116  to positive lead  34 . Output  102  is also connected via a resistor  114  to negative lead  36 . The anode of diode  108  is connected to an input  120  to which is conveyed, during operation, a square-wave signal u 1  ( FIG. 2A ) that is opposite in phase to a square-wave signal u 2  ( FIG. 2B ) that is conveyed to input  120 ′. 
   As  FIG. 2  shows, in this example signals u 1  and u 2  have an amplitude of U B /2 corresponding to the logical value “1”, or of &lt;0.4 V corresponding to the logical value “0”. Amplitude U B /2 is thus a linear function of operating voltage U B  (see  FIG. 3 ). 
   Negative input  124  of lower comparator  92 , like negative input  124 ′ of lower comparator  92 ′, is connected to node  52 , at which a potential of approximately (U B /2−0.7 V) is present, i.e. for example, for a voltage U B =20 V, a potential of approximately
 
(20/2−0.7)=9.3 V  (2).
 
   Positive input  126  of comparator  92  is connected via a resistor  128  to negative lead  36 , likewise to the anode of a diode  130  whose cathode is connected to terminal  120 . Input  126  is furthermore connected via a resistor  132  to output  100 . The latter is connected via a resistor  134 , a node  136 , and a resistor  138  to positive lead  34 . Node  136  is connected via a resistor  140  to gate G of upper transistor  70 . 
   Preferred values of the components in  FIG. 1  for a motor with U B =24 V (k=kilohm; R=resistor; C=capacitor): 
                                               Diodes 50, 62, 108, 108′, 130, 130′   bas16           Comparators 90, 90′, 92, 92′   LM2901           p-channel MOSFETs 70, 72   IRFR9024           n-channel MOSFETs 74, 76   IRFR024           C 96, 96′, 98, 98′      4 nF           R 44, 46     20 k           R 54, 58, 110, 110′, 128, 128′, 132, 132′     470 k           R 134, 134′    1.1 k           R 114, 114′, 136, 136′      3 k           R 116, 116′     510 ohm           R 112, 112′, 140, 140′     300 ohm           R 84, 86   0–0.1 ohm                        
Mode of Operation
 
   The description of the mode of operation makes reference only to the left half of  FIG. 1 , i.e. to the left half bridge. The right half is configured identically and therefore functions in the same way, but because signals u 1  and u 2  are opposite in phase, processes on the right side occur with a 180-degree phase shift from processes on the left side, as one skilled in the art of electrical engineering will readily understand. For example, when upper left transistor  70  is switched on, lower right transistor  76  is switched on approximately simultaneously; and when upper right transistor  72  is switched on, lower left transistor  74  is switched on approximately simultaneously. 
   At time t 1  in  FIG. 2A , signal u 1  at input  120  has a value of approx. 0.4 V, diode  130  becomes conductive, and the potential at positive input  126  therefore corresponds to the sum of that 0.4 V plus the voltage at diode  130  (approx. 0.7 V), i.e. approximately 1.1 V in total. This is lower than the potential at negative input  124 , which according to equations (1) and (2) is equal to (U B /2−0.7 V). Output  102  in lower comparator  92  is therefore connected internally to ground  36 , so that the UGS of transistor  74  is low and the latter is blocked. 
   The low potential at gate G of lower left transistor  74  is transferred through resistor  110  to positive input  106  of upper comparator  90 . This potential is lower than the potential (0.7 V) at negative input  104 , so that output  100  of upper comparator  90  is connected internally to ground  36 . By way of voltage divider  134  (1.1 k) and  138  (3 k), gate G of upper left transistor  70  acquires a potential of approx. 25% of operating voltage U B , so that at time t 1  transistor  70  is conductive while lower left transistor  74  blocks. 
   At time t 2  in  FIG. 2A , signal u 1  changes from 0.4 V to U B /2. This causes the previously conductive diode  130  to block, i.e. lower comparator  92  now serves to control the switching-on operation of lower left transistor  74  as a function of the gate potential of upper transistor  70 . Diode  108  now becomes conductive, and as a result a potential of approximately (U B /2−0.7 V) is present at positive input  106  of upper left comparator  90 . (The 0.7 V corresponds to the voltage at diode  108 , and U B /2 corresponds to the amplitude of signal u 1 .) Since this potential (U B /2−0.7 V) is higher than the reference potential of 0.7 V at negative input  104  of upper comparator  90 , the latter&#39;s output  100  becomes high-resistance, so that by way of the voltage divider made up of the four resistors  138 ,  134 ,  132 , and  128 , the potential at gate G of upper transistor  70  is pulled toward positive, and upper left transistor  70  consequently blocks. 
   The increase in the potential at gate G of upper transistor  70  is somewhat delayed by capacitor  96 , i.e. this capacitor determines the rate of increase. The parasitic capacitances in transistor  70  also contribute to this. The increase in the potential at output  100  is transferred via (identically sized) resistors  132  and  128  to positive input  126  of lower comparator  92 . Only when this potential has reached approximately twice the value (U B /2−0.7 V), i.e. when upper transistor  70  is safely blocked, is lower comparator  92  switched over to high resistance so that the voltage at gate G of lower left transistor  74  rises sufficiently that that transistor becomes conductive. Resistor  110  causes the potential at positive input  106  of upper comparator  90  to be raised so that the latter&#39;s output  100  remains at high resistance, and transistor  70  remains securely blocked. 
   It is thus evident that the change in potential at the gate of transistor  70  is transferred to positive input  126  of lower comparator  92 . The increase must be somewhat greater than twice (U B /2−0.7 V) because of voltage divider  128 ,  132 , and lower transistor  74  is switched on when that is the case. 
   In this case, upper transistor  70  is therefore switched off directly by means of signal u 1  via upper diode  108 , while lower diode  130  blocks and disconnects lower comparator  92  from signal u 1 , so that the latter component can delay the switching-on of transistor  74  until upper transistor  70  is safely blocked. 
   At time t 3  ( FIG. 2 ), signal u 1  changes from U B /2 to 0.4 V. As a result, diode  108  blocks, diode  130  becomes conductive, and positive input  126  of lower comparator  92  receives a potential of approximately 0.4 V+0.7 V=1.1 V. (The 0.7 V corresponds to the voltage at diode  130 .) Output  102  of lower comparator  92  is thereby connected internally to lead  36 , causing transistor  74  to block (after capacitor  98  discharges). 
   It should be noted here that the voltage values 0.4 V and 0.7 V are approximate values for a quantitative example, and that different values may occur in reality. 
   The change in the potential at output  102  is transferred through resistors  112 ,  110  to positive input  106  of upper comparator  90 . When the potential at input  106  becomes lower than the potential (0.7 V) at negative input  104 , output  100  of the upper comparator is then pulled down to the potential of negative lead  36 , so that the voltage USG between source and gate of upper transistor  70  rises correspondingly and the latter is switched on. 
   Upper transistor  70  is thus not switched on until the gate-source voltage UGS of lower transistor  74  has dropped to a value below the reference voltage of 0.7 V at positive input  104 , i.e. when lower transistor  74  is in the safe, high-resistance region. Resistors  128 ,  132  cause positive input  126  to receive a lower potential, so that the output of comparator  92  remains low-resistance and keeps transistor  74  blocked. At time t 3 , therefore, the previously conductive lower transistor  74  becomes blocked, and only when it is safely blocked is upper transistor  70  switched on. 
   It is evident that one of the two diodes  108 ,  130  serves respectively to block one of the two transistors  70 ,  74  immediately when signal u 1  changes, while the other diode blocks and thereby allows the comparator associated with it to operate as a comparator. This comparator compares the (variable) potential at its positive input with a reference voltage that, in this example, is equal to 0.7 V for upper comparator  90  and has a value of (U B /2−0.7 V) for lower comparator  92 ; in other words, for lower comparator  92 , this reference voltage is a function of voltage U B  and changes with it. This enables safe operation even when voltage U B  changes greatly during operation, e.g. as a result of charging or discharging of backup battery  40  that is depicted. 
     FIG. 3  shows an exemplifying embodiment for circuit  30  ( FIG. 1 ) in a simplified configuration in which the instant of commutation is not dependent on rotation speed. 
     FIG. 3  uses two comparators  150 ,  152  to whose inputs the output signal of Hall generator  28  is conveyed, as depicted, with reversed polarity. The two comparators  150 ,  152 , like comparators  90 ,  92 , are open-collector comparators, i.e. when the potential at the positive input of comparator  150  is higher than at the negative input, its output  32  is high-resistance; and when the potential at the positive input is lower than at the negative input, output  32  is connected internally to negative lead  36 , so that a potential of approx. +0.4V (with reference to negative lead  36 ) is obtained at output  32 . 
   Output  32  is connected via a resistor  154  to positive lead  34 , and via a resistor  156  to negative lead  36 . Resistors  154 ,  156  are identical in size, i.e. when output  32  is high-resistance, it acquires the potential UB/2 through resistors  154 ,  156 . 
   The same applies analogously to comparator  152 , whose output  33  is connected via a resistor  158  to positive lead  34  and via a resistor  160  to negative lead  36 . Resistors  158 ,  160  are also identical in size, so that once again a potential U B /2 exists at output  33  when that output is high-resistance, and a potential of 0.4 V when that output is connected internally to negative lead  36 . 
   The amplitude of signals u 1 , u 2  is thus directly proportional to voltage U B , which can vary within wide limits during operation. 
   Since the reference potential at node  52  of  FIG. 1  is also dependent on voltage U B , i.e. has the value (U B /2−0.7 V) in accordance with equation (1), the four comparators  90 ,  90 ′,  92 ,  92 ′ can reliably control full bridge circuit  78  even when voltage U B  changes. This is because the voltages that must be compared with one another using comparators  92 ,  92 ′ change in the same direction. Instead of U B /2 (i.e. a factor of 0.5), a different factor such as 0.6*U B  or 0.4*U B  could of course also be selected, in which case resistors  44 ,  46 ,  128 ,  132 ,  154 ,  156 ,  158 , and  160  would then need to be adapted accordingly. The approach using a factor of 0.5 is preferred, however, because in this case identical resistors with low tolerances can be used. 
   Many variants and modifications are, of course, possible within the scope of the present invention.