Abstract:
A tuner includes an RF input with a signal level controller such as an attenuator that is provided to attenuate a signal from the RF input. A tunable RF filter provides RF band limiting and determines simultaneously the AGC bandwidth. An amplifier with resistive feedback provides first and second outputs which, together, represent the incident power of the attenuated signal delivered by the amplifier, thus eliminating the need for a directional coupler and rendering this part of the tuner compatible with integration. A feedback circuit responds to the incident power of the attenuated signal by controlling the attenuation (AGC) provided by the signal level controller.

Description:
TECHNICAL FIELD OF THE INVENTION 
   The present invention relates to the control of the amount of attenuation within a tuner to prevent overload of the tuner&#39;s mixer or mixers. 
   BACKGROUND OF THE INVENTION 
   A tuner is provided in receivers such as radios and televisions to tune to a channel selected by a user. The tuner typically comprises a mixer that mixes the received RF signal with a local oscillator signal having a frequency corresponding to a selected channel. The output of the mixer is a predetermined intermediate frequency signal. In order to prevent the mixer from being overloaded by the received RF signal, an attenuator, such as a PIN diode attenuator, is provided between the RF input and the mixer. The attenuator is intended to ensure that the signal in the tuner is not so large so as to overload the mixer. 
     FIG. 1  shows an upconverting section  10  of a double conversion tuner without band limiting that provides an example of attenuation control in accordance with the prior art. The upconverting section  10  has an RF input  12  that receives an RF signal such as from an antenna. The RF signal is supplied to an attenuator  14  (such as a PIN diode attenuator) that is provided to attenuate the RF signal, and the RF signal at the output of the attenuator  14  is amplified by an RF amplifier  16 . The RF signal at the output of the RF amplifier  16  is supplied through a directional coupler  18  to a broadband filter  20  that is arranged to filter out signal components having frequencies outside of a selected range. For example, for television applications, the broadband filter  20  filters out signal components having frequencies outside of the frequency range of 50-800 MHz. The RF signal at the output of the broadband filter  20  is provided to the RF input of a mixer  22  which also receives a local oscillator signal LO from a local oscillator. The mixer  22  mixes the RF signal on its RF input with the local oscillator signal to produce an output IF signal. 
   The directional coupler  18  provides coupling to the incident voltage on the line between the RF amplifier  16  and the broadband filter  20  to an amplifier  24 , and this amplified voltage is provided to a detector  26 . In this way, the detector input level is not affected by input impedance variations of the broadband filter  20  which otherwise would cause attenuation variations. The detected signal at the output of the detector  26  is used to control the attenuator  14  such that, as the incident voltage at the output of the RF amplifier  16  becomes too large, the RF signal is attenuated to prevent overloading of the mixer  22 . 
   The problem with the upconverting section  10  is the cost of the directional coupler  18  and the technological incompatibility of the directional coupler  18  with the eventual tuner integration process. 
   SUMMARY OF THE INVENTION 
   In accordance with one aspect of the present invention, a tuner comprises an RF input, a signal level controller, a feedback amplifier, a mixer, and a feedback. The signal level controller is coupled to attenuate a signal from the RF input. The feedback amplifier is coupled to provide first and second outputs representing power of the attenuated signal. The mixer is coupled to mix at least one of the outputs of the amplifier with a local oscillator signal. The feedback is coupled to control the attenuation provided by the signal level controller in response to the power of the attenuated signal. 
   In accordance with another aspect of the present invention, a tuning method comprises the following: attenuating an RF signal received at an RF input; mixing the attenuated signal with a local oscillator signal to produce an intermediate frequency signal; developing a signal representing the power of the attenuated signal; and, controlling the attenuation of the RF signal in response to the signal representing the power of the attenuated signal so as to inhibit overloading of the mixer. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and other features and advantages of the present invention will become more apparent from a detailed consideration of the invention when taken in conjunction with the drawings in which: 
       FIG. 1  illustrates a prior art upconverting section of a double converting tuner without band limiting; 
       FIG. 2  illustrates an upconverting section of a double conversion tuner with RF band limiting in accordance with one embodiment of the present invention; 
       FIG. 3  illustrates an RF band limited double conversion tuner in accordance with another embodiment of the present invention; and, 
       FIG. 4  illustrates a conventional single conversion tuner improved in accordance with yet another embodiment of the present invention. 
   

   DETAILED DESCRIPTION 
   An upconverting section of a double conversion tuner  30 , as shown in  FIG. 2 , has an RF input  32  for receiving an RF signal such as from an antenna. The RF signal on the RF input  32  is supplied to an attenuator  34  (such as a PIN diode attenuator) which is provided to attenuate the RF signal. The RF signal at the output of the attenuator  34  is filtered by a tunable RF filter  36  and is amplified by an RF amplifier  38 . The RF signal at the output of the RF amplifier  38  is supplied to a broadband filter  40  that is arranged to filter out any signal components having frequencies outside of a selected range. For example, for television applications, the broadband filter  40  filters out signal components having frequencies outside of the frequency range of 50-800 MHz. The RF signal at the output of the broadband filter  40  is provided to the RF input of a mixer  42  which also receives a local oscillator signal LO from a local oscillator. The mixer  42  mixes the RF signal available at the output of the broadband filter  40  with the local oscillator signal LO to produce an output IF signal. 
   The RF amplifier  38  includes a transistor  44  having a gate  46  coupled to the output of the tunable RF filter  36 . The transistor  44  also has electrodes  48  and  50 . These electrodes may be drain and source electrodes in the case where the transistor  44  is a field effect transistor as shown in  FIG. 2 . However, the transistor  44  may be any other suitable type of an RF transistor. 
   The electrode  48 , which is the output of the RF amplifier  38 , is coupled to the broadband filter  40 . The electrode  48  is also coupled through a phase compensated divider  52  to an input  54  of an amplifier  56 . The input  54 , for example, may be a negative input of the amplifier  56 . The phase compensated divider  52  attenuates the voltage at the electrode  48  to a level comparable to the voltage at the electrode  50  and compensates for any phase shift caused by the RF amplifier  38 . The phase compensated divider  52  includes a resistor  58 , a capacitor  62 , and a resistor  64 . The resistor  58  and the capacitor  62  are coupled in parallel, and the parallel combination of the resistor  58  and the capacitor  62  couples the electrode  48  through a DC decoupling capacitor  60  to the input  54  of the amplifier  56 . The resistor  64  is coupled on one end to a junction of the resistor  58  and the DC decoupling capacitor  60  and on the other end to ground. 
   The electrode  50  of the transistor  44  is coupled through another DC decoupling capacitor  68  to an input  70  of the amplifier, and is also coupled to ground through a resistor  72  providing resistive series feedback and closing the DC circuit of the transistor  44  to ground. The input  70 , for example, may be a positive input of the amplifier  56 . A resistor  74  and a DC decoupling capacitor  76  are coupled in series between the gate  46  and the electrode  48  of the transistor  44  to provide parallel resistive feedback for the amplifier  38 . 
   The output of the amplifier  56  is coupled to a detector  78 , and the output of the detector  78  is coupled through a resistor  80  to the attenuator  34 . The junction of the resistor  80  and the attenuator  34  is coupled to ground through a capacitor  82  introducing thus the necessary delay into the AGC loop. The resistor  80 , which is in parallel with the attenuator control port input resistance and output resistance of the detector  78 , and the capacitor  82  determine a time constant for the automatic gain control. 
   The detected signal at the output of the detector  78  is used to control the attenuator  34  such that the RF signal is attenuated in order to prevent overloading of the mixer  42 . The voltage on the electrode  48  is directly the output voltage of the RF amplifier  38 , and the voltage on the electrode  50  is proportional to the current in the RF amplifier  38 . Therefore, the amplifier  56  receives two signals, a first signal representing the voltage of the attenuated signal (i.e., a properly scaled down voltage from the voltage at the output of the RF amplifier  38 ), and a second signal representing the current of the attenuated signal. These first and second signals provided by the RF amplifier  38  together closely represent the incident power at the output of the RF amplifier  38 . These first and second signals provided by the RF amplifier  38  are combined by the amplifier  56  and are then used to control the attenuator  34 . 
   Variations in the impedance of the broadband filter  40  are compensated because the first and second signals provided by the RF amplifier  38  complement each other as the impedance of the broadband filter  40  varies. For example, as the input impedance Z of the broadband filter  40  increases, the first signal on the electrode  48  increases while the second signal on the electrode  50  decreases, thereby stabilizing sufficiently the control of the attenuator  34 . 
   A double conversion tuner embodiment  100 , as shown in  FIG. 3 , has an RF input  102  for receiving an RF signal such as from an antenna. The RF signal is supplied to an attenuator  104  (such as a PIN diode attenuator) which is provided to attenuate the RF signal. The RF signal at the output of the attenuator  104  is filtered by a tunable RF filter  106  and is amplified by an RF amplifier  108 . The RF signal at the output of the RF amplifier  108  is supplied to a broadband filter  110  that is arranged to filter out signal components having frequencies outside of a selected broad frequency range. For example, for television applications, the broadband filter  110  filters out signal components having frequencies outside of the frequency range of 50-800 MHz. 
   The RF signal at the output of the broadband filter  110  is provided to the RF input of an upconverting first mixer  112  which also receives a local oscillator signal LO 1  from a local oscillator. The upconverting first mixer  112  mixes the RF signal from the broadband filter  110  with the local oscillator signal LO 1  to produce an initial intermediate frequency signal. 
   The initial intermediate frequency signal from the upconverting first mixer  112  is supplied to a wideband filter  114 . The purpose of the wideband filter  114  is mainly to minimize the effect of spurious frequencies generated by the mixing of the two local oscillator harmonics. The bandwidth of the wideband filter  114  has to be somewhat wider than the bandwidth of the tunable RF filter  106  for the AGC to function properly. The AGC frequency response has to be that of the RF filter  110  preceding the first mixer  112 . That would not be the case if the wideband filter  114  was narrower than the tunable RF filter  106 . 
   The signal at the output of the wideband filter  114  is amplified by an IF amplifier  116 . The signal at the output of the IF amplifier  116  is supplied to a narrowband IF filter  118  arranged to filter out signal components having frequencies outside of a selected range. For example, for television applications, the narrowband IF filter  118  ideally filters out all undesired signal components having frequencies outside of the frequency range of ±3 MHz from the first IF center frequency. This range depends on the technology that is used and the quality (cost) of the narrowband IF filter  118 . 
   The output of the narrowband IF filter  118  is provided to one input of a second downconverting mixer  120  which also receives a local oscillator signal LO 2  from a local oscillator. The second downconverting mixer  120  mixes the signal from the narrowband IF filter  118  with the local oscillator signal LO 2  to produce a final output IF signal usually around 44 MHz. 
   The IF amplifier  116  includes a transistor  122  having a gate  124  coupled to the output of the wideband filter  114 . The transistor  122  also has electrodes  126  and  128 . These electrodes may be drain and source electrodes in the case where the transistor  122  is a field effect transistor as shown in  FIG. 3 . However, the transistor  122  may be any other suitable type of transistor or amplifier. 
   The electrode  126  of the IF amplifier  116  is coupled to the input of the narrowband IF filter  118 . The electrode  126  is also coupled through a phase compensated divider  130  to an input  132  of an amplifier  134 . The input  132 , for example, may be a negative input of the amplifier  134 . The phase compensated divider  130  attenuates the voltage at the electrode  126  to a level comparable to the voltage at the electrode  128  and compensates for the phase shift caused by the IF amplifier  116 . The phase compensated divider  130  includes a resistor  136 , a capacitor  138 , a DC decoupling capacitor  140 , and a resistor  142 . The resistor  136  and the capacitor  138  are coupled in parallel, and the parallel combination of the resistor  136  and the capacitor  138  couples the electrode  126  through the DC decoupling capacitor  140  to the input  132  of the amplifier  134 . One end of the resistor  142  is coupled to the junction between the resistor  136  and the DC decoupling capacitor  140 , and the other end of the resistor  142  is coupled to ground. 
   The electrode  128  of the transistor  122  is coupled through another DC decoupling capacitor  144  to an input  146  of the amplifier  134 , and is also coupled to ground through a resistor  148  to provide resistive series feedback and to close the DC circuit of the transistor  122  to ground. The input  146 , for example, may be a positive input of the amplifier  134 . A resistor  152  and a DC decoupling capacitor  154  are coupled in series between the gate  124  and the electrode  126  of the transistor  122  to provide parallel resistive feedback for the IF amplifier  116 . 
   The output of amplifier  134  is coupled to a detector  156 , and the detector  156  is coupled through a resistor  158  to the attenuator  104 . The junction of the resistor  158  and the attenuator  104  is coupled to ground through a capacitor  160 . The resistor  158 , which is in parallel with the attenuator control port input resistance of the attenuator  104  and output resistance of the detector  156 , together with the capacitor  160  determine a time constant for the automatic gain control. 
   The detected signal at the output of the detector  156  is used to control the attenuator  104  such that the RF signal is attenuated in order to prevent overloading of the mixers  112  and  120 . The voltage on the electrode  126  directly is the output voltage of the IF amplifier  116 , and the voltage on the electrode  128  is proportional to the current in the IF amplifier  116 . Therefore, the amplifier  134  receives two signals, a first signal representing the voltage of the attenuated signal (i.e., a properly scaled down voltage from the voltage at the output of the IF amplifier  116 ), and a second signal representing the current of the attenuated signal. These first and second signals provided by the IF amplifier  116  complement each other and together represent closely the incident power at the output of the IF amplifier  116 . These first and second signals are combined by the amplifier  134  and used to control the attenuator  104 . 
   Variation in the input impedance of the narrowband IF filter  118  is compensated because the first and second signals provided by the IF amplifier  116  complement each other as the impedance of the narrowband IF filter  118  varies. For example, as the input impedance Z of the narrowband IF filter  118  increases, the signal on the electrode  126  increases while the signal on the electrode  128  decreases, thereby stabilizing control of the attenuator  104 . 
   The double conversion tuner embodiment  100 , as depicted in  FIG. 3 , has the advantage of providing a higher power level for the detector  156  compared with the double conversion tuner  30  of  FIG. 2 . Power sensing at a higher power level renders possibly the detector design easier and less costly. In addition, the AGC frequency spectrum to be amplified by the differential amplifier  134  and detected by the detector  156  is much narrower than the AGC spectrum width associated with the double conversion tuner  30 . The former AGC spectrum is a few tens of MHz (given by the RF filter bandwidth) centered around the first IF frequency while the latter is not converted to the first IF frequency. Also, the latter AGC spectrum has to accommodate the entire tuning range of the RF filter which is normally 50 to 800 MHz. The narrower the frequency spectrum to be amplified and detected by the differential amplifier  134  and the detector  156 , the easier is the implementation of both. 
   Certain modifications of the present invention have been discussed above. Other modifications will occur to those practicing in the art of the present invention. For example, a single conversion tuner  200  as depicted in  FIG. 4  could be economically designed in the conventional way with a varactor tuned single tuned first RF filter  202 , an AGC-able RF amplifier  204 , a varactor tuned double tuned second RF filter  206 , an active Gilbert cell mixer  208 , LO, a wideband IF filter  210 , a narrow band IF filter  212 , and a SAW driver  214  coupled to a SAW  216 . The new parts inserted into the otherwise standard single conversion tuner  200  would be an IF preamp stage  218  in front of the narrow band IF filter  212  to provide simultaneously the two taps for the AGC voltage and current signals. The two AGC signals are coupled to the symmetric input terminals of an amplifier  220  driving a detector  222 . 
   The wideband IF filter  210  may not be needed if the mixer isolation from the RF to the IF port and from the LO to the IF port is sufficiently large. The RF and AGC bandwidths are, for this design, basically identical. The bandwidth of the narrow band IF filter  212 , which is centered around 44 MHz, should be such as to protect the SAW driver  214  from strong adjacent channel interference. 
   The PIN attenuator used in the previous embodiments is now replaced with the AGC-able amplifier  204 , usually a dual gate MOSFET. 
   The bandwidth of the wideband filter  210 , if it has to be used, has to be somewhat wider than the RF bandwidth resulting from the first and second RF filters  202  and  206  combined. The signal driving the detector  222  is now band limited and centered around the IF frequency 44 MHz, which means that all the additional components become easy to design. 
   The embodiment as described in  FIG. 4  may also be expanded into a symmetric configuration starting with the active Gilbert cell mixer  208  up to the SAW driver  214  if it has to be implemented within an integrated circuit where most of the signal paths and signal processing blocks are designed as symmetric ones. 
   Accordingly, the description of the present invention is to be construed as illustrative only and is for the purpose of teaching those skilled in the art the best mode of carrying out the invention. The details may be varied substantially without departing from the spirit of the invention, and the exclusive use of all modifications which are within the scope of the appended claims is reserved.