Abstract:
A linear power control loop for a power amplifier is disclosed. Embodiments of the invention linearly control the power amplifier output, prevent the power amplifier from failure induced by excessive supply current by reducing power amplifier bias current, and provide a mechanism to detect whether the power amplifier is operating in a saturation condition.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    This invention relates generally to controlling the output power of a power amplifier. More particularly, the invention relates to a linear power control loop for controlling the output power of an amplifier contained in a portable communication handset. The invention also prevents an over-current condition and detects power amplifier saturation.  
           [0003]    2. Related Art  
           [0004]    With the increasing availability of efficient, low cost electronic modules, mobile communication systems are becoming more and more widespread. For example, there are many variations of communication schemes in which various frequencies, transmission schemes, modulation techniques and communication protocols are used to provide two-way voice and data communications in a handheld, telephone-like communication- handset, also referred to as a portable transceiver. The different modulation and transmission schemes each have advantages and disadvantages.  
           [0005]    As these mobile communication systems have been developed and deployed, many different standards have evolved, to which these systems must conform. For example, in the United States, many portable communications systems comply with the IS-136 standard, which requires the use of a particular modulation scheme and access format. In the case of IS-136, the modulation scheme is narrow band offset π/4 differential quadrature phase shift keying (π/4-DQPSK), and the access format is TDMA.  
           [0006]    In Europe, the global system for mobile communications (GSM) standard requires the use of the gaussian minimum shift keying (GMSK) modulation scheme in a narrow band TDMA access environment, which uses a constant envelope modulation methodology.  
           [0007]    Furthermore, in a typical GSM mobile communication system using narrow band TDMA technology, a GMSK modulation scheme supplies a low noise phase modulated (PM) transmit signal to a non-linear power amplifier directly from an oscillator. In such an arrangement, a highly efficient, non-linear power amplifier can be used thus allowing efficient modulation of the phase-modulated signal and minimizing power consumption. Because the modulated signal is supplied directly from an oscillator, the need for filtering, either before or after the power amplifier, is minimized. Further, the output in a GSM transceiver is a constant envelope (i.e., a non time-varying amplitude) modulation signal.  
           [0008]    Regardless of the type of modulation methodology employed, the output power supplied by the power amplifier must be controlled to provide the most efficient power level for the conditions under which the communication handset is operating. For example, in the GSM communication system, the power amplifier transmits in bursts and must be able to control the ramp-up of the transmit power as well as have a high degree of control over the output power level over a wide power range. This power control is typically performed using a feedback loop in which a portion of the signal output from the power amplifier is compared with a reference signal and the resulting error signal is fed back to the control input of the power amplifier.  
           [0009]    In some other communication systems, the output power is controlled by a signal from the base station with which the portable transceiver is communicating. Typically, in such an arrangement, the base station simply sends a signal to the portable transceiver instructing the portable transceiver to increase or decrease power. In such systems, there is no specific power requirement, just the command to either increase or decrease power output.  
           [0010]    Regardless of the type of power control employed, the output of the power amplifier is preferably controlled in precise steps. For communication handsets that use a bipolar transistor power amplifier, the output of the power amplifier is controlled by a control signal that is applied to the base terminal of the final stage (if multiple amplifier stages are employed) of the power amplifier. This is commonly referred to as the “base bias current.” 
           [0011]    As the conditions (e.g., temperature, battery voltage, antenna impedance, etc.) under which the communication handset operates vary, the power control loop acts to maintain the output power of the power amplifier constant by adjusting the base bias current. Increasing the base bias current generally causes the output of the power amplifier to increase.  
           [0012]    While a conventional power control loop provides some control over the power output, some problems may arise. For example, if the base bias current increases past a certain level, the power amplifier is susceptible to failure. This can happen, for example, if the impedance of the antenna abruptly changes due to, for example, a change in the position of the portable transceiver relative to nearby reflective surfaces.  
           [0013]    Another problem with a conventional power control loop is that the ratio of the base bias current to the output power characteristic is non-linear. At higher power levels, the level of the base bias control current must be disproportionately (i.e., non-linearly) raised to achieve a commensurate increase (in dB) in output power. This causes the “loop gain” of the power control loop to decrease at higher output power levels, which lengthens the response time of the power control loop. This manifests as an inability to quickly shut off the transmitter, which is a problem in systems such as GSM in which a burst transmission methodology demands fast power ramp-up and ramp-down times.  
           [0014]    Linearizing the power control loop has been previously attempted by inserting a “shaper” stage into the power control loop. The shaper is a filter that is designed to exhibit a non-linear gain that has the inverse characteristic of the ratio of the base bias current to the output power. Unfortunately, it is difficult to obtain an exactly inverse characteristic, and the loop gain of the power control loop still decreases at higher output power, causing the response time of the power control loop to decrease.  
           [0015]    Previous power control systems have protected the power amplifier by using various types of power amplifier saturation detection methodologies. Conventional saturation detectors sense the decrease in loop gain by observing a large error signal in the power control loop. Unfortunately, these systems require additional circuitry to reduce the base bias current when saturation is detected. For example, a logical “saturation detect” signal may be directed to a microprocessor or digital signal processor (DSP) in the communication handset. The microprocessor or DSP then directs the power control loop to reduce its target value until the saturation detect signal is cleared. A disadvantage of this type of system is that a single threshold is chosen to determine when the power amplifier has become, or is becoming, saturated. This is problematic because the power amplifier will not operate above this level, while the true saturation point of the power amplifier may be dependent on temperature and other variables that change over time.  
           [0016]    Another conventional power control system measures only the base bias current. This protects the power amplifier from burning out, but measuring the base bias current provides only an indirect indication of the output power of the power amplifier.  
           [0017]    Therefore it would be desirable to provide a power control loop for a power amplifier that exhibits linear response and that includes saturation detection and over-current protection.  
         SUMMARY  
         [0018]    Embodiments of the invention include a linear power control loop for a power amplifier. Embodiments of the invention linearly control the power amplifier output, prevent the power amplifier from failure induced by excessive supply current by reducing power amplifier bias current, and provide a mechanism to detect whether the power amplifier is operating in a saturation condition. In one embodiment, the invention is a linear power control loop for a power amplifier, comprising a detector for providing a direct current (DC) baseband signal representing an output of a power amplifier, a comparator for comparing the DC baseband signal to a reference signal and generating an error signal, a filter, and a power amplifier control current driver for linearly controlling the output of the power amplifier.  
           [0019]    Related methods of operation and computer readable media are also provided. Other systems, methods, features, and advantages of the invention will be or become apparent to one with skill in the art upon examination of the following figures and detailed description. It is intended that all such additional systems, methods, features, and advantages be included within this description, be within the scope of the invention, and be protected by the accompanying claims. 
       
    
    
     BRIEF DESCRIPTION OF THE FIGURES  
       [0020]    The invention can be better understood with reference to the following figures. The components within the figures are not necessarily to scale, emphasis instead being placed upon clearly illustrating the principles of the invention. Moreover, in the figures, like reference numerals designate corresponding parts throughout the different views.  
         [0021]    [0021]FIG. 1 is a block diagram illustrating a simplified portable transceiver.  
         [0022]    [0022]FIG. 2 is a block diagram illustrating the power control element of FIG. 1.  
         [0023]    [0023]FIG. 3 is a schematic diagram illustrating a first embodiment of the PA control current driver of FIG. 2.  
         [0024]    [0024]FIG. 4 is a schematic diagram illustrating a first alternative embodiment of the PA control current driver of FIG. 2.  
         [0025]    [0025]FIG. 5 is a, schematic diagram illustrating a second alternative embodiment of the PA control current driver of FIG. 2.  
         [0026]    [0026]FIG. 6 is a graphical illustration showing the relationship between the power output (Pout) of the power amplifier in dBM, and the control voltage, Vctrl, on connection  172  of FIG. 2. 
     
    
     DETAILED DESCRIPTION  
       [0027]    Although described with particular reference to a portable transceiver, the linear power control loop having over-current protection and saturation detection (referred to below as the “linear power control loop”) can be implemented in any system that uses a bipolar power amplifier having bias current control.  
         [0028]    The linear power control loop can be implemented in software, hardware, or a combination of software and hardware. In a preferred embodiment, the linear power control loop may be implemented in hardware. The hardware of the invention can be implemented using specialized hardware elements and logic. If portions are implemented in software, the software portion can be stored in a memory and be executed by a suitable instruction execution system (microprocessor). The hardware implementation of the linear power control loop can include any or a combination of the following technologies, which are all well known in the art: a discrete logic circuit(s) having logic gates for implementing logic functions upon data signals, an application specific integrated circuit having appropriate logic gates, a programmable gate array(s) (PGA), a field programmable gate array (FPGA), etc.  
         [0029]    The software of the linear power control loop comprises an ordered listing of executable instructions for implementing logical functions, and can be embodied in any computer-readable medium for use by or in connection with an instruction execution system, apparatus, or device, such as a computer-based system, processor-containing system, or other system that can fetch the instructions from the instruction execution system, apparatus, or device and execute the instructions.  
         [0030]    In the context of this document, a “computer-readable medium” can be any means that can contain, store, communicate, propagate, or transport the program for use by or in connection with the instruction execution system, apparatus, or device. The computer readable medium can be, for example but not limited to, an electronic, magnetic, optical, electromagnetic, infrared, or semiconductor system, apparatus, device, or propagation medium. More specific examples (a non-exhaustive list) of the computer-readable medium would include the following: an electrical connection (electronic) having one or more wires, a portable computer diskette (magnetic), a random access memory (RAM), a read-only memory (ROM), an erasable programmable read-only memory (EPROM or Flash memory) (magnetic), an optical fiber (optical), and a portable compact disc read-only memory (CDROM) (optical). Note that the computer-readable medium could even be paper or another suitable medium upon which the program is printed, as the program can be electronically captured, via for instance optical scanning of the paper or other medium, then compiled, interpreted or otherwise processed in a suitable manner if necessary, and then stored in a computer memory.  
         [0031]    [0031]FIG. 1 is a block diagram illustrating a simplified portable transceiver  100 . Portable transceiver  100  includes speaker  102 , display  104 , keyboard  106 , and microphone  108 , all connected to baseband subsystem  110 . In a particular embodiment, portable transceiver  100  can be, for example but not limited to, a portable telecommunication handset such as a mobile/cellular-type telephone. Speaker  102  and display  104  receive signals from baseband subsystem  110  via connections  112  and  114 , respectively, as known to those skilled in the art. Similarly, keyboard  106  and microphone  108  supply signals to baseband subsystem  110  via connections  116  and  118 , respectively. Baseband subsystem  110  includes microprocessor (μP)  120 , memory  122 , analog circuitry  124 , and digital signal processor (DSP)  126  in communication via bus  128 . Bus  128 , although shown as a single bus, may be implemented using multiple busses connected as necessary among the subsystems within baseband subsystem  110 . Microprocessor  120  and memory  122  provide the signal timing, processing and storage functions for portable transceiver  100 . Analog circuitry  124  provides the analog processing functions for the signals within baseband subsystem  110 . Baseband subsystem  110  provides control signals to radio frequency (RF) subsystem  130  via connection  132 . Although shown as a single connection  132 , the control signals may originate from DSP  126  or from microprocessor  120 , and are supplied to a variety of points within RF subsystem  130 . It should be noted that, for simplicity, only the basic components of portable transceiver  100  are illustrated herein.  
         [0032]    Baseband subsystem  110  also includes analog-to-digital converter (ADC)  134  and digital-to-analog converters (DACs)  136  and  138 . Although DACs  136  and  138  are illustrated as two separate devices, it is understood that a single digital-to-analog converter may be used that performs the function of DACs  136  and  138 . ADC  134 , DAC  136  and DAC  138  also communicate with microprocessor  120 , memory  122 , analog circuitry  124  and DSP  126  via bus  128 . DAC  136  converts the digital communication information within baseband subsystem  110  into an analog signal for transmission to RF subsystem  130  via connection  140 . DAC  138  provides a reference voltage power level signal to power control element  200  via connection  144 . Connection  140 , while shown as two directed arrows, includes the information that is to be transmitted by RF subsystem  130  after conversion from the digital domain to the analog domain. As will be described in detail below, in one embodiment of the invention, the power control element  200  provides a saturation detection signal (e.g.,sat_detect) via connection  198  to the baseband subsystem  110 .  
         [0033]    RF subsystem  130  includes modulator  146 , which, after receiving a frequency reference signal, also called a “local oscillator” signal, or “LO,” from synthesizer  148  via connection  150 , modulates the received analog information and provides a modulated signal via connection  152  to upconverter  154 . In a constant envelope modulation methodology, the modulated transmit signal generally includes only phase information. Upconverter  154  also receives a frequency reference signal from synthesizer  148  via connection  156 . Synthesizer  148  determines the appropriate frequency to which the upconverter  154  upconverts the modulated signal on connection  152 .  
         [0034]    Upconverter  154  supplies the modulated signal via connection  158  to power amplifier  160 . Power amplifier  160  amplifies the modulated signal on connection  158  to the appropriate power level for transmission via connection  162  to antenna  164 . Illustratively, switch  166  controls whether the amplified signal on connection  162  is transferred to antenna  164  or whether a received signal from antenna  164  is supplied to filter  168 . The operation of switch  166  is controlled by a control signal from baseband subsystem  110  via connection  132 . Alternatively, the switch  166  may be replaced by a filter pair (e.g., a duplexer) that allows simultaneous passage of both transmit signals and receive signals, as known in the art.  
         [0035]    A portion of the amplified transmit signal energy on connection  162  is supplied via connection  170  to power control element  200 . The power control element  200  generally forms a closed power control feedback loop to control the output power of power amplifier  160  and may also supply a power control feedback signal via connection  172 . In accordance with an aspect of the invention, the power control element is linear in that it supplies a power amplifier control signal via connection  172  that is linearly proportional to the power output (measured in dBm) of the power amplifier  160 . Generally, the power amplifier  160  has a control characteristic that is substantially linear with respect to the input control current (I CTRL  IN)/power output (dBm OUT) until the power amplifier enters saturation. The power control element  200  is generally linear with respect to power input (dBm IN)/output control current (I CTRL  OUT). Therefore, negating power in dBm, the power control element  200  has a logarithmic characteristic, while the power amplifier  160  has an anti-logarithmic characteristic. When taken together, the characteristics of the power amplifier  160  and the power control element  200  result in a linear power control loop. In accordance with alternative embodiments of the invention, the power control element  200  also provides protection against the power amplifier entering an over-current state, which could cause the power amplifier  160  to fail. The power control element  200  also provides a saturation detection function that can determine whether the power amplifier  160  is operating in a saturated mode, and if so detected, inform the baseband subsystem  110 , via connection  198 , which in turn may reduce the output of the power amplifier via the power amplifier control signal on connection  172  by a signal sent over connection  144  to the power control element  200 .  
         [0036]    A signal received by antenna  164  will be directed to receive filter  168 . Receive filter  168  will filter the received signal and supply the filtered signal on connection  174  to low noise amplifier (LNA)  176 . Receive filter  168  is a band pass filter, which passes all channels of the particular cellular system in which the portable transceiver  100  is operating. As an example, for a 900 MHz GSM system, receive filter  168  would pass all frequencies from 935.2 MHz to 959.8 MHz, covering all 124 contiguous channels of 200 kHz each. The purpose of this filter is to reject all frequencies outside the desired region. LNA  176  amplifies the very weak signal on connection  174  to a level at which downconverter  178  can translate the signal from the transmitted frequency to an IF frequency. Alternatively, the functionality of LNA  176  and downconverter  178  can be accomplished using other elements, such as, for example but not limited to, a low noise block downconverter (LNB).  
         [0037]    Downconverter  178  receives a frequency reference signal, also called a “local oscillator” signal, or “LO”, from synthesizer  148 , via connection  180 , which signal instructs the downconverter  178  as to the proper frequency to which to downconvert the signal received from LNA  176  via connection  182 . The downconverted frequency is called the intermediate frequency or IF. Downconverter  178  sends the downconverted signal via connection  184  to channel filter  186 , also called the “IF filter.” Channel filter  186  filters the downconverted signal and supplies it via connection  188  to amplifier  190 . The channel filter  186  selects the one desired channel and rejects all others. Using the GSM system as an example, only one of the  124  contiguous channels is actually to be received. After all channels are passed by receive filter  168  and downconverted in frequency by downconverter  178 , only the one desired channel will appear precisely at the center frequency of channel filter  186 . The synthesizer  148 , by controlling the local oscillator frequency supplied on connection  180  to downconverter  178 , determines the selected channel. Amplifier  190  amplifies the received signal and supplies the amplified signal via connection  192  to demodulator  194 . Demodulator  194  recovers the transmitted analog information and supplies a signal representing this information via connection  196  to ADC  134 . ADC  134  converts these analog signals to a digital signal at baseband frequency and transfers the signal via bus  128  to DSP  126  for further processing. As an alternative, the downconverted carrier frequency (IF frequency) at connection  184  may be 0 Hz, in which case the receiver is referred to as a “direct conversion receiver.” In such a case, the channel filter  186  is implemented as a low pass filter, and the demodulator  194  may be omitted.  
         [0038]    [0038]FIG. 2 is a block diagram illustrating the power control element  200  of FIG. 1. For simplicity, the function of the modulator  146  and the upconverter  154  of FIG. 1 are illustrated in FIG. 2 using oscillator  202 . Oscillator  202 , which may be a voltage controlled oscillator (VCO), supplies a low-noise modulated signal (i.e, a signal with very low out-of-band noise) via connection  158  to the power amplifier  160 . By using an oscillator  202  to supply a low-noise modulated signal to power amplifier  160 , the need for filtering before and after the power amp  160  may be reduced or eliminated.  
         [0039]    A portion of the output power present on connection  162  is diverted by coupler  210  via connection  170  to a logarithmic (log) detector  212 . The log detector  212  receives the RF signal on connection  170  and provides, on connection  214 , a direct current (DC) baseband signal representing the level of the RF power signal present on connection  170 . The output of the log detector  212  is supplied on connection  214  to a comparator  218 . A reference voltage power control signal from the DAC  138  of FIG. 1 is supplied via connection  144  to the comparator  218 . The comparator  218  compares the signal level on connection  214  with the signal level on connection  144  and provides an error signal representing the difference on connection  222 .  
         [0040]    The error signal on connection  222  is supplied to a low pass filter (LPF)  226 . The LPF  226  acts as an integrator, integrating the error signal on connection  222  to provide a control voltage signal on connection  232 . The control voltage signal on connection  232  represents the difference between the output power measured from the power amplifier  160  and the desired output level supplied on connection  144 , integrated over time such that the result is essentially proportional to the desired power output of the power amplifier in dBm.  
         [0041]    The control voltage signal on connection  232  is supplied to the power amplifier (PA) control current driver  300 . The power amplifier control current driver  300 , as will be described in detail below, provides a linear control signal via connection  172  to the control input of the power amplifier  160 . The power amplifier control current driver  300  receives the control voltage signal on connection  232  and supplies a control current signal on connection  172 . The power amplifier control current driver also receives a battery voltage (V+) on connection  234  and provides a power amplifier supply voltage to the power amplifier  160  via connection  242 .  
         [0042]    [0042]FIG. 3 is a schematic diagram illustrating a first embodiment  310  of the power amplifier control current driver  300  of FIG. 2. The power amplifier control current driver  310  is supplied by a voltage source  312  having one terminal coupled to ground  316  via connection  314 . The voltage source  312  can be, for example, a battery located in a communication handset that supplies a DC voltage of approximately 3.8 volts (V). Alternatively, the voltage source can be other devices, such as, for example, a fuel cell.  
         [0043]    The voltage source  312  is coupled to the power amplifier control current driver  310  via connection  234 . The voltage level on connection  234  can be referred to as the “battery voltage.” In this embodiment, the power amplifier control current driver  310  includes a transistor  320 . The transistor  320  includes a collector terminal  324 , a base terminal  332 , and an emitter terminal  326 . The transistor can be referred to as a power control transistor. Preferably, the transistor  320  is a bipolar junction transistor (BJT), but a field effect transistor (FET) could also be suitable. The battery voltage on connection  234  is coupled to the collector terminal  324  of the transistor  320 . The operation power supplied to the power amplifier  160  (FIG. 2) may come from the same voltage source  312 , via connection  242 , or it may come from another source.  
         [0044]    The connection  232  is the output of the low pass filter  226  (FIG. 2) and is coupled to a resistor  328 . The resistor  328  is also coupled to the base terminal  332  of the transistor  320 . Another resistor  334  is coupled to the base terminal  332  of transistor  320 . The resistor  328  preferably has a value of 50 Ω and the resistor  334  preferably has a value of 100 Ω. The emitter terminal  326  of the transistor  320  is also coupled to one terminal of the resistor  334 . This node forms the output  172  of the power control element  200  (FIG. 2). A power amplifier control current signal is supplied over connection  172  to the power amplifier  160 . In a heterojunction bipolar transistor (HBT) power amplifier, the power amplifier control current signal is supplied to the base terminal (not shown, but coupled to connection  172 ) of the power amplifier  160  (FIG. 2).  
         [0045]    In accordance with an aspect of the invention, the power amplifier control current signal provided on connection  172  increases proportionally and substantially linearly with the RF power output of the power amplifier  160  (FIG. 2). The power amplifier control current signal on connection  172  directly affects the power gain and power output of the power amplifier  160 . However, for increasing levels of output power, a disproportionately large increase in the voltage on the control input (connection  172 ) of the power amplifier  160  is needed to achieve the desired current output from the power amplifier  160 . This is illustrated below in the graph  600  shown in FIG. 6, where the voltage on the control input  172  of the power amplifier  160  is referred to as “Vctrl” and the RF output power of the power amplifier  160  is referred to as “Pout” (dBm) and is plotted against Vctrl. The graph  600  includes a trace  610 , which includes a linear region  615  and a saturated region  620 .  
         [0046]    As shown in FIG. 6, for an exemplary bipolar transistor, for voltage levels above approximately 1.1V, the output of the power amplifier  160  increases disproportionately in response to the voltage Vctrl, eventually driving the power amplifier into saturation. The line  630  represents an ideal linear relationship between power output and control voltage. Without the power amplifier control current driver  300 , the control voltage, Vctrl, would have to be supplied by connection  232  (FIG. 2). Since the control voltage signal on connection  232  is the integrated error signal of the control loop, it is desired that the RF power output of power amplifier  160  be proportionally and linearly related to this voltage. Therefore, the function of the power amplifier control current driver  300  is to drive additional current, as needed, to effectively linearize the relationship between the RF power output of the power amplifier  160  and the control voltage signal on connection  232 .  
         [0047]    For example, as the power amplifier  160  demands more current from the supply via connection  242  (in order to provide more output power), the current input on connection  232  (i.e., due to the control voltage signal provided by the low pass filter  226  of FIG. 2), continues to increase. As the voltage (and current) on connection  232  continues to increase, the current through resistors  328  and  334  increases and-therefore, the voltage across resistor  334  increases. The voltage across resistor  334  is the base-to-emitter voltage (V BE ) of transistor  320 . As V BE  of transistor  320  increases, the transistor  320  gradually turns on and supplies an increasing amount of current from its collector terminal  326 . This current from transistor  320  constitutes a supplemental current, which is added to the power amplifier control current signal on connection  172  (FIG. 3). The increased current on connection  172  causes the power amplifier  160  (FIG. 2) to consume more power from the voltage source  312  via connection  242  and therefore the RF power gain of the power amplifier  160  will increase. Because the transistor  320  adds the supplemental current from the emitter terminal  326  to the power amplifier control current signal via connection  172 , it makes the RF power output of the power amplifier  160  substantially proportional and linear to the voltage on connection  232 .  
         [0048]    The transistor  320  begins to turn on at a predetermined level, which is determined by the value of the resistor  334 . Assuming that no supplemental current is added by the transistor  320 , there will be some current flowing through resistor  334  that, when driven into the power amplifier control input on connection  172 , corresponds to a voltage at the power amplifier control input  172 . The transistor  320  should begin to turn on when the current through the resistor  334  causes the voltage across the resistor  334  to reach approximately 0.7V. When the transistor  320  turns on its current increases as the input voltage (and current) on connection  232  increases. Therefore, as the power amplifier  160  demands more supply current, the power amplifier gets the current via connection  242  while the input voltage on connection  232  need not increase disproportionately with respect to the desired RF output power of the power amplifier  160 . Rather, the level of the control voltage signal on connection  232  need only increase approximately linearly with respect to the output of the power amplifier  160  (FIG. 2).  
         [0049]    [0049]FIG. 4 is a schematic diagram illustrating a first alternative embodiment  400  of the PA control current driver  300  of FIG. 2. FIG. 4 contains many elements that are similar in function to those described above in FIG. 3. The elements in FIG. 4 that are similar to the elements in FIG. 3 are similarly labeled. For example, the transistor  320  of FIG. 3 is labeled transistor  420  in FIG. 4. The transistor  420  in FIG. 4 performs the same function as the transistor  320  in FIG. 3. Accordingly, the components in FIG. 4 that were previously described in FIG. 3 will not again be described in detail.  
         [0050]    In FIG. 4, an additional transistor  430 , resistor  466  and resistor  486 , all coupled in parallel with the transistor  320 , resistor  428  and resistor  434  is illustrated. The transistor  430  and related circuitry can be used in an implementation in which the portable communication device, in which the PA control current driver is located, is capable of transmitting multiple communication methodologies. For example, the input on connection  232  can be directed by switch  488  onto connection  478  for a first modulation type while the input on connection  232  may be directed by switch  488  onto connection  482  if modulation of a different type is possible using the portable communication handset in which the power amplifier control current driver is located. Similarly, the switch  492  directs the output of either the transistor  420  or the transistor  430  onto connection  172 . The switches  488  and  492  can be controlled by, for example, a control signal from the baseband subsystem  110  via connection  132  (FIG. 1).  
         [0051]    In response to the power amplifier  160  (FIG. 2) demanding increased current so that it can provide higher output power, the current at the collector terminal  424  of transistor  420  will continue to increase in response to the power demand from the power amplifier  160 . Unfortunately, the current at the collector terminal  424  of transistor  420  will continue to increase until the power amplifier  160  consequently draws too much current from its supply, and fails catastrophically. This is a condition is sometimes referred to as “burn-out.” 
         [0052]    In the embodiment illustrated in FIG. 4, the power amplifier control current driver  400  includes an over-current protection element  444 . The over-current protection element  444  comprises a transistor  440  configured as a diode, and a transistor  450 . The transistor  440  includes an emitter terminal  446 , which is coupled to battery voltage on connection  418  via resistor  442 . Preferably, the resistor  442  has a value of approximately 50 Ω. The collector terminal and the base terminal of the transistor  440  are shorted via connection  448 , thus forming a diode.  
         [0053]    The shorted collector and base terminal of transistor  440  provides a constant base bias voltage to the base terminal  452  of transistor  450 . The transistor  450  includes an emitter terminal  454  coupled to battery voltage on connection  418  through the resistor  422  and a collector terminal  456  coupled to the collector terminal  424  of the transistor  420 . As the current on the collector terminal  424  of transistor  420  continues to rise, it causes the current drawn from the power amplifier  160  (FIG. 2) via connection  242  to eventually reach a predetermined level at which the transistor  450  begins to shut off. In this manner, the power amplifier is prevented from failing due to an over-current condition.  
         [0054]    To determine the predetermined level at which the transistor  450  begins to shut off, a maximum power amplifier supply current is determined. The maximum power amplifier supply current is chosen to prevent the power amplifier  160  from burning out. This current causes a certain voltage drop across the resistor  422 . The value of the resistor  442  is then chosen to have the same voltage drop, but with a significantly lower current. For example, if the value of the resistor  422  is 50 mΩ and if the value of the resistor  442  is 50Ω, the current consumed by resistor  442  to obtain the desired bias current is {fraction (1/1000)} of the maximum power amplifier supply current.  
         [0055]    For example, the voltage at the base terminal  452  of the transistor  450  is at a fixed level. Accordingly, when the current drawn from the power amplifier  160  (FIG. 2) via connection  242  reaches a predetermined level, the voltage drop across the emitter terminal  454  and the base terminal  452  of the transistor  450  causes the transistor  450  to begin turning off. The transistor  450  is biased using the transistor  440  configured as a diode to maintain a constant voltage on the base terminal  452  of transistor  450 . The emitter terminal  454  is supplied from the voltage source  412  via connection  418  through the resistor  422 , through which also passes the power amplifier supply current via connection  242 . The value of the resistor  422  is preferably 50 mΩ. Thus, the transistor  450  compares the voltage drop across the 50 mΩ resistor  422  to a reference voltage on connection  452  and begins to turn off when the current drawn via connection  242  causes a sufficient voltage increase across the 50 mΩ resistor  422 .  
         [0056]    As the transistor  450  begins to turn off, the transistor  420  (or the transistor  430  if used) can no longer supply as much supplemental control current to connection  172 . Therefore, the power amplifier control signal on connection  172  is reduced, thereby preventing the power amplifier  160  (FIG. 2) from entering an over-current condition and possibly failing.  
         [0057]    [0057]FIG. 5 is a schematic diagram illustrating a second alternative embodiment  500  of the PA control current driver  300  of FIG. 2. The elements in FIG. 5 that have been previously discussed are numbered similarly to those in FIGS. 3 and 4 and will not again be described in detail. FIG. 5 includes a saturation detection transistor  560  having an emitter terminal  538  coupled to battery voltage on connection  518  via resistor  536 . Preferably, the value of the resistor  536  is 100 Ω. The collector terminal  596  of the transistor  560  is coupled through a resistor  602  to ground  606  via connection  604 . Preferably, the value of the resistor  602  is 1 KΩ. The base terminal  594  of the transistor  560  is coupled to the collector terminal  556  of the transistor  550 .  
         [0058]    During normal operation of the power amplifier  160  (i.e., when the power amplifier  160  is not in danger of an over-current condition), the transistor  550  is on, and the voltage present at the collector terminal  556  of transistor  550  is at a level that is slightly below (approximately 100 millivolts (mV)) the level of the supply voltage of the voltage source  512  on connection  518 . Therefore, the voltage drop from the base  594  to the emitter  538  of the transistor  560  is small, causing transistor  560  to be shut off. Accordingly, the voltage at the collector terminal  596  of the transistor  560  is close to zero, as this point is pulled down through the resistor  602 .  
         [0059]    When the PA supply current on connection  242  increases until it reaches a pre-defined over-current condition, the over-current protection is provided by the transistor  550  being turned off (as described in FIG. 4), and the voltage on the base terminal  594  of the transistor  560  will decrease until the transistor  560  will turn on, thereby causing the voltage across the resistor  602  to be pulled up to the voltage of the voltage source  512  through the resistor  536 . The resistor  602  is coupled to the connection  198 . When the transistor  560  turns on, the voltage level on the connection  198  goes up to essentially the same voltage as the battery voltage. This can be used as a saturation detection (sat_detect) signal and is supplied to the baseband subsystem  110  of FIG. 1 via connection  198 . In this manner, the baseband subsystem  110  can act to reduce the current supplied to the power amplifier  160  (FIG. 2) by requesting a lower power level via the control signal on connection  144  (FIG. 1).  
         [0060]    The saturation detection transistor  560  operates as described because the collector terminal  556  of the transistor  550  provides a convenient signal for the detection of the power amplifier  160  (FIG. 2) reaching its maximum operating point. As the power amplifier  160  reaches its maximum operating point, the voltage at the collector terminal  556  of the transistor  550  begins to drop. The transistor  560  is connected such that it is off when the power amplifier  160  is providing low power and turns on when the power amplifier  160  reaches maximum power, or any other desired threshold as determined by the value of the resistor  536 . Therefore, when the power amplifier  160  reaches maximum power, the sat_detect signal is generated by the transistor  560  and supplied to the baseband subsystem  110  (FIG. 1).  
         [0061]    [0061]FIG. 6 is a graphical illustration  600  showing the relationship between the power output (Pout) of the power amplifier  160  in dBm and the control voltage, Vctrl, on connection  172  (FIG. 2). The power output of the power amplifier is represented on the vertical axis and the control voltage (V ctrl ) on connection  172  is represented on the horizontal axis. As shown, the power output of the power amplifier  160  remains substantially linear for control voltages between approximately 0.9V through 1.1V.  
         [0062]    While various embodiments of the invention have been described, it will be apparent to those of ordinary skill in the art that many more embodiments and implementations are possible that are within the scope of this invention. Accordingly, the invention is not to be restricted except in light of the following claims and their equivalents.