Abstract:
An apparatus and method for measuring the breakpoint of a response curve representing the voltage output of an image array having an extended dynamic range. By flooding a light-opaque pixel with a charge and then applying an intermediate reset voltage to the pixel, the signal is read from the pixel and stored. The full reset voltage is applied to the pixel, and then the signal in the pixel is read and stored. The voltage output difference is the difference between the first and second stored signal. The voltage output difference is then used to determine the voltage of the knee point. Further, a conventional saturated pixel can be reset with an intermediate reset just prior to readout. The resulting signal can then be used to determine the voltage of the knee point.

Description:
The present invention relates generally to imager array operation, and more specifically, to circuits and methods for changing the dynamic range of an imager array. 
   BACKGROUND OF THE INVENTION 
   There is a current interest in CMOS active pixel imagers for use as low cost imaging devices.  FIG. 1  shows a conventional CMOS active pixel sensor imager array circuit  100  including a pixel array  110  and associated circuitry. Circuit  100  includes an array  110  of pixels  150  and a row decoding/controlling circuit  130  and a column address/decoding/readout circuit  120  which provide timing and control signals to enable reading out of signals stored in the pixels  150  in a manner commonly known. The array  110  has columns  149  and rows  147  of pixels  150 . Exemplary arrays  110  have dimensions of M times N pixels  150 , with the size of the array  110  depending on a particular application. The imager  100  is read out a row  147  at a time using a column parallel readout architecture. The row circuit  130  selects a particular row  147  of pixels  150  in the array  110  by controlling the operation of row addressing and row drivers (not shown) within the row circuit  130 . Charge integration signals stored in the selected row of pixels  150  are provided on the column lines  170  to a column circuit  120 , in a manner described below, and sampled and stored in column buffer circuits  151 . The pair of signals corresponding to the read out reset signal and integrated charge signal (e.g., Vrst, Vsig) are provided by the column buffer circuits  151  to a differential amplifier circuit  154 . The differential amplifier circuit  154  provides the differential output of the pixel signals (e.g., Vrst, Vsig) to the analog-to-digital (ADC) circuit  156 . The ADC circuit  156  provides a digital value representing the signal of the pixel  150  output to associated circuitry  199 . Associated circuitry  199  is representative of the many circuits that receive the input from the ADC circuit  156  and perform operations on the input. For example, associated circuitry  199 , may store, transfer to a bus/memory, and perform linear/non-linear operations on the signal input. 
   A pixel  150  of the CMOS active pixel sensor imager array  100  is shown in greater detail in  FIG. 2 . Pixel  150  can have one or more active transistors within the pixel unit cell, can be made compatible with CMOS technologies, and promises higher readout rates compared to passive pixel sensors. The  FIG. 2  pixel  150  is a 3T APS, where the 3T is commonly used in the art to designate use of three transistors to operate the pixel. A 3T pixel has a photodiode  162 , a reset transistor  184 , a source follower transistor  186 , and a row select transistor  188 . It should be understood that while  FIG. 2  shows the circuitry for operation of a single pixel  150 , in practical use there will be an M times N array of identical pixels  150  arranged in rows and columns with the pixels  150  of the array accessed using row and column select circuitry, as described above. 
   The photodiode  162  converts incident photons to electrons which collect at node A. A source follower transistor  186  has its gate connected to node A and amplifies the signal appearing at node A. When a particular row-containing cell  150  is selected by a row selection transistor  188 , the signal amplified by transistor  186  is passed on a column line  170  to the readout circuitry. The photodiode  162  accumulates a photo-generated charge in a doped region of the substrate. It should be understood that the pixel  150  might include a photogate or other photoconversion device, in lieu of a photodiode, for producing photo-generated charge. 
   A reset voltage source Vrst, typically Vaa, on line  195  is selectively coupled through reset transistor  184  to node A. The row select control line  160  is coupled to all of the pixels  150  of the same row of the array. Voltage source Vaa is coupled to a source following transistor  186  and its output is selectively coupled to a column line  170  through row select transistor  188 . Although not shown in  FIG. 1 , column line  170  is coupled to all of the pixels of the same column of the array and typically has a current sink at its lower end. The gate of row select transistor  188  is coupled to row select control line  160 . 
   The gate of reset transistor  184  is coupled to reset control circuit  180  through reset control line  191 . Reset control circuit  180  serves to control the reset operation in which Vrst is coupled to node A. Reset control circuit  180  may provide a plurality of control signals to the reset transistor  184 , e.g., the reset control circuit  180  provides a full and an intermediate reset signal to reset transistor  184 . Reset control circuit  180  is mutually coupled to the reset transistor  184  of each pixel  150  in the row  147 . Each row  147  of pixels  150  has an associated reset control circuit  180 . 
   As known in the art, a value is read from pixel  150  in a two-step process. During a charge integration period, the photodiode  162  converts photons to electrons which collect at the node A. The charges at node A are amplified by source follower transistor  186  and selectively passed to column line  170  by row access transistor  188 . During a reset period, node A is reset by turning on reset transistor  184  and the reset voltage is applied to node A and read out to column line  170  by the source follower transistor  186  through the activated row select transistor  188 . As a result, the two different values—the reset voltage Vrst and the image signal voltage Vsig—are readout from the pixel and sent by the column line  170  to the readout circuitry where each is sampled and held for further processing as known in the art. 
     FIG. 3  more clearly shows the column buffer circuit  151  of  FIG. 1  that is capable of sampling and holding and then providing two sampled values, e.g., Vsig and Vrst values, for subsequent use by a down stream circuit ( FIG. 1 ). As seen in  FIG. 3  the column line  170  is switchably coupled through SH_R switch  310  to the first side of capacitor  318 . The second side of capacitor  318  is switchably coupled through switch  326  to a downstream circuit. The column line  170  is also switchably coupled through SH_S switch  310  to the first side of capacitor  320 . The second side of capacitor  320  is switchably coupled through switch  328  to a downstream circuit. The first side of capacitor  318  is switchably coupled through switch  313  to the first side of capacitor  320 . A clamp voltage Vcl is switchably coupled through switch  315  to the second side of capacitor  318 . A clamp voltage Vcl is also switchably coupled through switch  317  to the second side of capacitor  320 . 
   The conventional CMOS imager array  100  ( FIG. 1 ) has a limited dynamic range and is prone to over-saturation from receiving a very high light intensity. As is known in the art, the dynamic range of a pixel array can be increased by implementation of an extended dynamic range (XDR) technique, where a modified reset signal is applied to the gate of the reset transistor. The modified reset signal, referred to as an intermediate reset value, has a different, typically smaller, amplitude than the “full reset” signal asserted at the start of each integration period. The intermediate reset value will affect only those cells on which a strong light signal is incident. Only if a photocurrent has reduced the voltage across a cell&#39;s photodiode to below a certain level at the time the intermediate reset value is asserted, then the voltage across the photodiode will be pulled up to Vdac=Vxdr−Vth, where Vxdr is the voltage, i.e., of the intermediate voltage, on the reset line  191  ( FIG. 2 ) and Vth is the threshold voltage of the reset transistor (e.g., reset transistor  184  of  FIG. 2 ). In accordance with the XDR technique, during each integration period, the intermediate reset value is applied to the gate of each of the reset transistors in the row. 
     FIG. 4  depicts a Vadc output of an upstream pixel from an imager system when the XDR technique is applied. By applying an intermediate reset value during the integration period, the response curve of the image sensor is converted from a linear curve to a piecewise linear curve, as illustrated in  FIG. 4 . 
   The vertical axis of  FIG. 4  represents digital data output from circuit  156  ( FIG. 1 ), i.e., an ADC amplifier  156 , as a result of a read of an upstream pixel  150  of the imager  100  ( FIG. 1 ). The Vzero  411  axis represents a zero (“0”) Vadc output. The Vadc_break  413  axis represents a Vadc output of the kneepoint (e.g., the break point). The Vadc_max  417  axis represents a maximum Vadc output of a pixel  150 , e.g., the saturation point. The horizontal axis in  FIG. 4  represents incident light intensity on the cell during the integration period. The Lzero  401  axis represents a zero (“0”) light intensity output. The Lbreak  403  axis represents a light intensity output of the kneepoint. The Lno_XDR  405  axis represents a maximum light intensity output of a pixel  150  if an XDR technique is not applied, e.g., the saturation point of the pixel without an extended dynamic range. The LXDR  407  axis represents a maximum light intensity output of a pixel  150  if an XDR technique is applied, e.g., the saturation point of the pixel with an extended dynamic range. 
   Curve  410  represents the response curve of pixel  150  ( FIG. 1 ), indicating the range of detectable incident light intensity (from Lzero  401  to Lno_XDR  405 ) corresponding to the full range of the output of ADC amplifier  156  ( FIG. 1 ), from Vzero  411  to Vadc_max  417 , when no intermediate reset signal is asserted during the integration period. Curve  420  of  FIG. 4  represents the response curve of the pixel  150 , indicating the extended range of detectable incident light intensity (from Lzero  401  to LXDR  407 , where intensity LXDR  407  is greater than Lno_XDR) corresponding to the full range of the output of ADC amplifier  156 , when an intermediate reset signal is asserted during the integration period. The break point  430  represents the change in the response of pixel  150  to light intensity after an intermediate pulse signal is provided, i.e., after the XDR technique is applied. At breakpoint  430 , the response curve changes from the non-extended range light intensity curve  410  to the extended range light intensity curve  420 . 
     FIG. 4  illustrates how the XDR technique extends the dynamic range of the image sensor, namely by increasing the maximum detectable light intensity from Lno_XDR  405  to LXDR  407 . The coordinates of the breakpoint ( 403 ,  413 ) depend on the photodiode voltage immediately after an XDR reset (Vdac=Vxdr−Vth) and the time at which the XDR reset is performed during the integration period. It is possible to perform two or more intermediate resets in a single integration period, which results in a piecewise linear sensor response curve having N+1 linear sections, and N breakpoints (one breakpoint for each of N intermediate resets), and can (in some cases) increase the dynamic range beyond that achievable with only one intermediate reset per integration period. 
   When implementing the XDR technique, the voltage supplied to the reset line  191  ( FIG. 2 ) can be generated using a DAC (digital-to-analog converter). The imager  100  ( FIG. 1 ) can be programmed with a desired intermediate voltage level and the time at which each XDR reset is performed. 
     FIG. 5  illustrates how the XDR technique affects the digital output from an ADC circuit  156  ( FIG. 1 ) of an upstream pixel cell  150  over time. The vertical axis of  FIG. 5  represents voltage data output from circuit  156  ( FIG. 1 ), i.e., an ADC amplifier  156 , as a result of a read of an upstream pixel  150  of the circuit  100  ( FIG. 1 ). 
   The Vzero  511  axis represents a zero (“0”) Vadc output. The Vadc_break  515  axis represents a Vadc output of the kneepoint. The Vadc_max  517  axis represents a maximum Vadc output of a pixel  150 , e.g., the saturation point. 
   The horizontal axis in  FIG. 5  represents time. The Tzero  501  axis represents time zero (“0”). The time Trst  503  axis represents the time of the kneepoint, e.g., the time that the intermediate reset voltage is applied. The time Tint  507  axis represents the total integration time of a pixel  150 , regardless whether a XDR technique is applied, e.g., the saturation point of the pixel with an extended dynamic range. 
   Curve  550  represents the response curve of pixel  150  ( FIG. 1 ) when the XDR mode is disabled and the intermediate reset is not applied. The slope of curve  550  (i.e., dV/dt) is proportional to the photocurrent of pixel  150  and the intensity of light incident on pixel  150 . As drawn, curve  550  corresponds to the maximum light intensity that can be sensed by pixel  150  without saturation when XDR is off, Imax_noxdr. Any curve with a steeper slope, like curve  510 , will cause the output of pixel  150  to saturate at Vadc_max  517 . 
   Curves  510  and  520  represent the response curve of pixel  150  when the XDR mode is enabled. Any pixel  150  receiving more light than that represented by curve  540  will be affected by the intermediate reset. While a fill reset would cause integration to restart from Vzero, the intermediate reset restarts integration from Vadc_intrst  513  and continues for a duration of (Tint−Trst). During this second period of integration the maximum light intensity that will not cause pixel saturation is represented by the slope of curve  520 , Imax_xdr. 
   Curve  510  represents the response curve of pixel  150  ( FIG. 1 ) indicating the range of time (from Tzero  501  to Trst  503 ) corresponding to the full range of the output of the ADC amplifier  156  (from Vzero  511  to Tint  517 ), when no intermediate reset signal is asserted during the integration period. Curve  520  of  FIG. 5  represents the response curve of pixel  150 , indicating the extended range of time, Tint (from Trst  503  to Tint  507 , where Tint  507  is greater than Trst  503 ) corresponding to the full range of output of the ADC amplifier  156  when an intermediate reset signal is asserted during the integration period at the breakpoint time period Trst  503 . 
   Point  530  shows that, upon application of the intermediate reset signal, the response of pixel  150  to photon integration changes. At time period  503 , the response curve changes from the non-extended range voltage output curve  510  to the extended range voltage output curve  520 . The Vadc_instr  513  represents the change in Vrst that is provided, i.e., the full reset value offset by the intermediate reset value. 
     FIG. 5  illustrates how the XDR technique extends the dynamic range of the image sensor, namely by increasing the maximum amount of exposure for the pixel  150  without causing the signal from the pixel  150  being over saturated. 
   A pixel array with an XDR system can be programmed to provide a desired kneepoint by specifying the desired Trst (i.e., intermediate reset time) and intermediate reset voltage Vrst. However, within sensor circuitry are inherent sources of variation affecting absolute signal magnitudes. Although a XDR system is designed to provide intermediate and full reset voltages, process variation can affect the provided voltage, resulting in actual intermediate and full reset voltages, which are different from the desired intermediate and full reset voltages. Therefore, the desired knee point may differ from the actual knee point. 
   The knee point corresponds to the time when the intermediate reset voltage is applied and the voltage output from the pixel at the break point, i.e., Vadc_break. The knee point can be determined by first determining the difference between the full reset voltage and the intermediate reset, i.e., ΔVrst. Since values of Trst and Tint are pre-determined, Vadc_break can be calculated once ΔVrst is determined, where:
 
 Vadc _break=( Tint/Trst )*Δ Vrst   (1).
 
   However, it is unknown how to determine the actual difference between the full reset voltage and the intermediate reset, i.e., ΔVrst. Consequently, it is not been known how to determine the actual knee point of the response curve. Thus, it would be desirable to be able to determine the actual knee point of the response curve. 
   SUMMARY OF THE INVENTION 
   The present invention provides a method and apparatus for determining the knee point of a response curve of an image array applying an extended dynamic range technique. 
   In a first exemplary embodiment, the knee point is determined by first flooding the pixel artificially by coupling the light sensitive node pixel to ground. Then an intermediate reset voltage is applied to the pixel and the signal in the pixel is sampled and held. Subsequently, a full reset voltage is applied to the pixel, and the signal in the pixel sampled and held. Using the two signals from the pixel, the difference between the intermediate and full reset value is determined. The pixel is either light opaque or shielded from incident light. 
   In another exemplary embodiment, the knee point is determined by sampling and storing the signal values from the image array without applying the XDR technique. The signal values of saturated pixels are recorded. Then, the image array is sampled and stored applying the XDR technique, where the intermediate reset voltage is applied at a time Trst, where Trst is substantially equivalent to Tint. The difference between the intermediate and full reset value is determined by comparing the difference between the signal values of saturated pixels from the initial readout of the pixel array and the signal values of the corresponding pixels from the subsequent readout of the pixel array. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and other features and advantages of the invention will be more readily understood from the following detailed description of the invention which is provided in connection with the accompanying drawings, in which: 
       FIG. 1  is a block diagram of a conventional CMOS image array and associated circuitry; 
       FIG. 2  is a block diagram of a conventional three transistor pixel cell of the  FIG. 1  CMOS image array; 
       FIG. 3  is a block diagram of a conventional column buffer circuit of the  FIG. 1  CMOS image array; 
       FIG. 4  is a graph depicting a conventional knee-point curve comparing Vadc with light intensity; 
       FIG. 5  is a graph depicting a conventional curve comparing Vadc over time; 
       FIG. 6  is a block diagram of a CMOS image array and associated circuitry in accordance with an exemplary embodiment of the invention; 
       FIG. 7  is block diagram showing a portion of the  FIG. 6  diagram in greater detail; 
       FIG. 8  is a simplified timing diagram associated with the operation of the circuitry of  FIG. 6 ; 
       FIG. 9  is a simplified timing diagram associated with the operation of the circuitry of  FIG. 1  in accordance with another exemplary embodiment of the invention; 
       FIG. 10  is a graph depicting a knee-point curve comparing Vadc over time in accordance with a second embodiment of the invention; 
       FIG. 11  is a block diagram representation of an imaging device in accordance with an exemplary embodiment of the invention; and 
       FIG. 12  is a block diagram representation of a processor-based system incorporating an imaging device in accordance with an exemplary embodiment of the invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 6  depicts a CMOS active pixel sensor imager array circuit  600  including associated circuitry according to an exemplary embodiment of the present invention. Circuit  600  differs from circuit  100  in that circuit  600  includes a power supply circuit  690 . Power supply circuit  690  mutually couples line  695  either to a reset voltage Vaa or to ground. Power supply circuit  690  is mutually coupled to the reset transistor  184  ( FIG. 2 ) of each pixel  150  in the row  147 . The row  147  of pixels  150  is representative of many rows  147  of pixels  150  in the array  110  (similar to  FIG. 1 ) though only one row  147  of pixels  150  is shown. Each row  147  of pixels  150  has an associated power supply circuit  690 . The pixel  150  is shielded from incident light. 
   Within the power supply circuit  690 , line  695  is mutually coupled to a first source/drain of respective transistors  692 ,  694 . The other source/drain of transistor  692  is coupled to a Vaa voltage. The other source/drain of transistor  694  is coupled to ground. A signal line  697  is coupled to the gate of transistor  694  and inversely coupled to the gate of transistor  692 ; thus, only one of transistors  692 ,  694  is closed (i.e., conductive) at a time. Depending on the signal carried on line  697 , line  695  is either coupled to Vaa or ground. For example, if the signal carried on line  697  is logic high, then transistor  692  is open and non-conductive, and transistor  694  is closed and conductive. Alternatively, if the signal carried on line  697  is logic low, then transistor  694  is open and non-conductive, and transistor  692  is closed and conductive. Therefore, power supply circuit  690  either provides a Vaa voltage to the row  147  of pixels  150 , or connects row  147  to ground. 
     FIG. 7  is block diagram showing a portion of the  FIG. 6  diagram in greater detail.  FIG. 7  depicts pixel  150  being coupled to the power supply circuit  690 . More specifically, power supply circuit  690  is coupled through line  695  to the reset transistor  184  and source follower transistor  186 .  FIG. 7 , also depicts pixel  150  being coupled to the reset control circuit  690  through reset control line  695 . 
   Turning to  FIG. 8 , the basic operation of the circuits of  FIGS. 3 ,  6 , and  7  is now described with reference to sampling and storing a set of signals from a pixel  150 , where the two signals that are applied are the full reset value and the intermediate reset value. With reference to  FIGS. 3 ,  6 , and  7 , the control signals are shown over defined time intervals  990 ,  992 ,  994 , and  996 . 
   The first time interval  990  is a flood time interval. During time interval  990 , the light sensitive node of pixel  150  is coupled to ground, thereby flooding pixel  150  with a negative charge to force saturation, e.g., removing any stored signal from the pixel  150 . The second time interval  992  is a Trst time interval. During Trst interval  992 , the intermediate reset signal is provided to the pixel  150 , a corresponding voltage is coupled to the pixel  150 , and the pixel  150  is reset with respect to the intermediate reset value. The third time interval  994  is a Tint 2  time interval. During the Tint 2  time interval  994 , the voltage value stored at node A ( FIG. 7 ) is sampled and stored in capacitor  320  of column circuit buffer  151  ( FIG. 3 ). The combined time intervals  992  and  994  comprise the Tint time interval  998 . The fourth time interval  996  is a post-Tint time interval. During post Tint time interval  996 , a full value reset signal is provided to the pixel  150  and a corresponding voltage is coupled to the pixel  150 , and the pixel  150  is charged with respect to the full reset value. Also, during the post Tint time interval  996 , the voltage value stored at node A ( FIG. 7 ) is sampled and stored in capacitor  318  of column circuit buffer  151  ( FIG. 3 ). 
   Throughout  FIG. 8 . a logic high signal indicates that the corresponding transistor of  FIGS. 3 ,  6 , and  7  is closed (conductive), while a logic low signal indicates that the corresponding transistor of  FIGS. 3 ,  6 , and  7  is open (non-conductive). 
   RST/GRND signal  901  corresponds to the logic level of transistor  694  and the inverse logic level of transistor  692  in power supply circuit  690  ( FIG. 6 ). RST signal  903  corresponds to the logic level of reset transistor  184  in pixel  150  ( FIG. 7 ). ROW signal  907  corresponds to the logic level of row select transistor  188  in pixel  150  ( FIG. 7 ). SHR signal  909  corresponds to the logic level of sampling switch  312  in column buffer circuit  151  ( FIG. 3 ). SHS signal  911  corresponds to the logic level of sampling switch  310  in column buffer circuit  151  ( FIG. 3 ). 
   During time interval  902 , transistor  692  opens and transistor  694  closes to couple line  695  to ground. During a time interval  904 , transistor  184  is provided a full reset value and is closed to couple pixel  150  to ground; thereby, flooding pixel  150  with a negative charge. Time interval  902  begins before time interval  904  begins, and time interval  902  ends after time interval  904  ends. After time interval  902  ends transistor  692  is closed and transistor  694  opens to couple line  695  to Vaa. 
   During time interval  905 , transistor  184  is provided an intermediate reset value and partially closes to couple pixel  150  to Vaa through line  695 ; thereby, providing an intermediate Vaa voltage to pixel  150 . During time interval  908 , row transistor  188  closes to couple node A of pixel  150  to column buffer circuit  151  through column line  170 . Time  908  begins before the earlier of either time intervals  910 ,  912  begins, and ends after the later of time intervals  910 ,  912  ends. Time interval  908  begins after time interval  905  ends. 
   During time interval  912  sampling switch  312  closes, coupling pixel  150  through line  170  to charge capacitor  320  with the value stored in pixel  150 . Time interval  912  begins after time interval  908  begins and ends before time interval  906  begins. During time interval  906 , transistor  184  is provided a full reset value and closes to couple pixel  150  to Vaa through line  695 , thereby providing a full Vaa voltage to pixel  150 . Time interval  906  begins after time interval  912  ends and ends before time interval  910  begins. 
   During time interval  910 , sampling switch  310  closes, coupling pixel  150  through line  170  to charge capacitor  318  with the value stored in pixel  150 ; where the value stored in pixel  150  is substantially equivalent to the full reset value. Time interval  910  begins after time interval  906  ends and ends before time interval  908  ends. Thus, the set of signals, i.e., the intermediate reset value and the full reset value, from the pixel  150  are stored in the column buffer  151 . ( FIG. 3 ) 
   After the set of signals are stored in the column buffer circuit  151 , they are provided to the differential amplifier  154 , which in turn, provides the outputted, differentiated signals to the ADC amplifier  156 . ( FIG. 6 ) The ADC amplifier  156  then provides an output voltage signal. The output voltage signal represents the difference between the provided full reset voltage and the provided intermediate reset voltage. Given that voltage at the break point is related to the integration time of the pixel and the difference between the full reset voltage and the intermediate reset voltage, the voltage at the break point is computed as: 
   
     
       
         
           
             
               
                 
                   
                     
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   Since Tint and Trst are known and ΔVrst is provided by the ADC amplifier, the kneepoint can be determined. 
   In another aspect of the present invention, pixel  150  is located in a redundant area of the array  110 . 
   In another exemplary embodiment of the present invention, the intermediate reset voltage is determined by using conventional circuitry and changing the intermediate reset time, and without the need for special light-opaque pixels and power supply circuit.  FIG. 9  is a timing diagram that depicts the operation of circuit  100  according to this embodiment of the invention. The timing diagram of  FIG. 9  differs from the timing diagram of  FIG. 8  in several respects. First, the circuit  100  in the timing diagram of  FIG. 9 , the image array  110  is read in two segments, first without the XDR technique enabled during time period  1001 , and second with the XDR technique enabled during time period  1002 . Second, circuit  100  does not have a power supply circuit  690 , and therefore no control signals are provided for that circuit. Third, the time when the intermediate reset voltage is provided during the XDR technique, e.g., Trst, is substantially close to the time of the integration, e.g., Tint. 
   The second segment  1002  of the readout from the image array is depicted in  FIG. 10  where the knee-point curve is shown comparing Vadc over time. As seen in  FIG. 10 , Trst  803  is closer to Tint  507  so that the difference between Trst  803  and Tint  507  is minimized. 
   As indicated above, in this exemplary embodiment of the invention, the kneepoint is determined using a two-step process. In the first step of the process, an XDR technique is disabled and not employed, and the CMOS image array  100  ( FIG. 1 ) is read in the conventional fashion and processed. The values, and possibly locations, of saturated pixels are recorded. For example, a pixel indicates a measured light intensity, after being processed by an ADC amplifier on a graduated scale from 0 to 1,023, where 1,023 is a saturated pixel. Then, a pixel having a measured light intensity level close in value to 1,023 is recorded, e.g., for example, those pixels having a measured light intensity level greater than 1,018. If the measurement of the pixels indicates that there are no pixels that are saturated, then the exposure is adjusted to force a number of pixels to have a value indicating that they are saturated. 
   In the second step of the process during time period  1002  ( FIG. 9 ), the image array is read out again. During this readout, an XDR technique is enabled. As indicated above in this exemplary embodiment, the Trst occurs just briefly before Tint. In a preferred embodiment, the Trst is substantially equal to Tint. The values are read from the pixels of the array and processed. The post-ADC processed signals taken in the second step of the process that correspond to the pixels identified in the first step of the process as saturated pixels in the first are compared to determined the difference in voltage output from these pixels in the first and second step of the process. The result of the comparison provides the ΔVrst. As described above with respect to Equation (2), once ΔVrst is determined, then Vadc_break can be calculated. 
     FIG. 11  illustrates a block diagram of an exemplary imager device  1108  that may be used in accordance with an embodiment of the invention. Imager  1108  has a pixel array  1100  and row lines are selectively activated by a row driver  1110  in response to row address decoder  1120 . A column driver  1160  and column address decoder  1170  are also included. The imager device  1108  is operated by the timing and control circuit  1150 , which controls address decoders  1120 ,  1170 . The control circuit  1150  also controls the row and column driver circuitry  1110 ,  1160 . A sample and hold circuit  1161  associated with the column driver  1160  reads a pixel reset signal (V rst ) and a pixel image signal (V sig ) for the selected pixels. A differential signal (V rst −V sig ) is produced by differential amplifier  1162  for each pixel. The differential signal is digitized by analog-to-digital converter  1175  (ADC). The analog-to-digital converter  1175  supplies the digitized pixel signals to an image processor  1180 , which forms and outputs a digital image. 
   The method and apparatus aspects of the invention are embodied in an imager device  1240  shown in  FIG. 12 , which provides an image output signal. The imager device  1240  may be, for example, the imager device  1108  of  FIG. 11 . The image output signal can also be applied to a processor system  1200 , also illustrated in  FIG. 12 . A processor based system, such as a computer system, for example, generally comprises a central processing unit (CPU)  1210 , for example, a microprocessor, that communicates with one or more input/output (I/O) devices  1250  over a bus  1270 . The CPU  1210  also exchanges data with random access memory (RAM)  1260  over bus  1270 , typically through a memory controller. The processor system may also include peripheral devices such as a floppy disk drive  1220  and a compact disk (CD) ROM drive  1230  which also communicate with CPU  1210  over the bus  1270 . Imager device  1240  is coupled to the processor system and includes a pixel storage and readout circuit as described along with respect to  FIG. 6 . 
   While the invention has been described and illustrated with reference to specific exemplary embodiments, it should be understood that many modifications and substitutions can be made without departing from the spirit and scope of the invention. For example, although described with reference to a 3T pixel, the invention is not so limited. Further, although described with reference to CMOS active pixel arrays the invention is not so limited. Accordingly, the invention is not to be considered as limited by the foregoing description but is only limited by the scope of the claims.