Abstract:
A method of spatial control of a phased array system having a plurality of antenna elements is provided. The method includes providing a baseband signal, baseband phase shifting the baseband signal to provide a plurality of baseband shifted signals for controlling phase of each of the plurality of antenna elements, upconverting each of the baseband shifted signals to a radio frequency signal, and applying each of the radio frequency signals to the plurality of antenna elements to thereby provide for spatial control of the phased array system. A hardware architecture for a phased array system is also provided.

Description:
FIELD OF THE INVENTION 
     The present invention relates to a phased array antenna system, and more particularly to a method and system for spatial control of a phased array antenna system. 
     BACKGROUND OF THE INVENTION 
     Phased array antenna systems have many applications in wireless, especially MIMO (multiple inputs and multiple outputs) communication. By using multiple antennas to transmit and receive the signal, the transmit rate is pushed closer towards the channel capacity limit while simultaneously improving security. 
     Another application of such a system is in sensor array networks where information from a single sensor can be collected or transmitted to a specific receiver by steering the antenna in the right direction. Since the transmitted signal is steered to a specific receiver and nulled in other directions, the security of the signal is improved. A phased array antenna system can be utilized by the military to transmit and receive secure information. A phased array antenna system also has applications in mobile LANs, adaptive dynamic array processing for antennas and automotive radars for collision control, path/lane control, etc. 
     However, problems remain with phased array antenna systems. Of particular concern is accurate adjustability of the phase and amplitude characteristics for each element of a phased array. Therefore what is needed is an improved method and system for spatial control of a phased array antenna system. 
     BRIEF SUMMARY OF THE INVENTION 
     According to one aspect of the present invention, a method of spatial control of a phased array system having a plurality of antenna elements is provided. The method includes providing a baseband signal, baseband phase shifting the baseband signal to provide a plurality of baseband shifted signals for controlling phase of each of the plurality of antenna elements, upconverting each of the baseband shifted signals to a radio frequency signal, and applying each of the radio frequency signals to the plurality of antenna elements to thereby provide for spatial control of the phased array system. 
     According to another aspect of the present invention, a phased array system is provided. The phased array system includes a plurality of integrated antenna elements, a phase adjusting circuit comprising active phase shifters adapted to provide baseband phase shifts in a baseband signal, and an upconverter circuit operatively connected between the phase adjusting circuit and the plurality of integrated antenna elements and adapted to upconvert the baseband signal to a radio frequency signal. 
     According to another aspect of the present invention, a phased array system, includes a plurality of integrated antenna elements, a phase detecting circuit adapted to detect baseband phase shifts in a signal, and a downconverter circuit operatively connected between the phase detecting circuit and the plurality of integrated antenna elements and adapted to downconvert the signal. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates a Microstrip and a Dipole Array at an operating frequency of 2.425 GHz. The Dipole Array has a length of 1203 mils and spaced 1155 mils apart. Microstrip Array has a length of 1270 mils, width of 1453 mil, probe feed at position 730 mils by 420 mils on each patch, and patches are spaced 310 mils apart. 
         FIG. 2  provides graphs showing a comparison of experimental versus standard and active impedance corrected results for a microstrip and Dipole antenna, respectively [5]. 
         FIG. 3  is a polar E-field plot for the field pattern verses theoretical plot including mutual coupling effects, shifted at two different angles using the automatic phase shifter shown in  FIG. 4 . 
         FIG. 4  illustrates a variable delay frequency synthesizer at 2.425 GHz. 
         FIG. 5A  is a block diagram of a phase locked loop with a digital divider. 
         FIG. 5B  is a block diagram of a phase locked loop which uses a phase aid. 
         FIG. 6  is the signal shifted at baseband operation and at RF at two different types of delays. 
         FIG. 7  is an experimental phase error for a phase locked loop using digital dividers with a reference frequency of 3.125 MHz and 2.425 GHz. 
         FIG. 8  is a simulated phase error plot for the PLL unit step phase response [5]. 
         FIG. 9  is a quadrature phase shifting keying and quadrature amplitude modulation communication scheme. 
         FIG. 10  is a simulated phase error plot for the PLL unit step phase response. 
         FIG. 11  is a block diagram of one embodiment of a system. 
         FIG. 12  is a block diagram of a receive system. 
         FIG. 13  is a block diagram of a transmitting/receiving system. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     The present phased array antennas are useful for many types of wireless communications. To facilitate description of phased array antennas of the present invention, a discussion regarding theory and modeling is provided, hardware designs are shown, and testing setup and results are provided. 
     Theory and Modeling 
     Standard beam formations can be written in terms of the element factor and the array factor as shown
 
 E   array   =E   element   ·E   arrayfactor   (1)
 
     This assumption ignores the fact that there is mutual coupling. Eq. 2 and 3 show the standard field pattern for a two element dipole and microstrip array as shown in  FIG. 1 . 
                     E   dipole     H   -   plane       =       E   0     ⁢     cos   ⁡     [       (         k   0     ⁢   d   ⁢           ⁢   sin   ⁢           ⁢   ϕ     +   β     )     /   2     ]                 (   2   )                 E   microstrip     E   -   plane       =         E   0     ⁡     [       (       k   0     ⁢     h   /   2       )     ⁢   cos   ⁢           ⁢   ϕ     ]       ⁢     cos   ⁡     [       (       k   0     ⁢     L   /   2       )     ⁢   sin   ⁢           ⁢   ϕ     ]       ⁢     cos   ⁡     [       (         k   0     ⁢   d   ⁢           ⁢   sin   ⁢           ⁢   ϕ     +   β     )     /   2     ]                 (   3   )               
where k 0 =2π/λ 0 ,β are the free space wave number and phase difference of the excitation at the antenna, respectively [1].
 
Scanning Angle
 
     An important aspect of a phased array antenna is the ability to steer the main beam in the direction containing the line of sight, thus reducing multi-path fading, which can be described by the Rician distribution [2]. As shown in [3], the main beam of an antenna can be steered by controlling the phases of the current on the elements as shown 
                       I   v       I   0       =              I   v       I   0            ⁢     ⅇ       -   j     ⁢           ⁢     k   0     ⁢         r   →     v     ·     p   →                     (   4   )               
where,
 
 {right arrow over (p)} =sin φ 0  cos φ 0   â   x +sin θ 0   â   y +cos θ 0   â   z ,  (5)
 
and (θ 0 ,φ 0 ) are the scanning angles in spherical coordinates. It can then be shown in [3] that grating lobes can appear at angles
 
                       sin   ⁡     (     θ   gl     )       =       sin   ⁢           ⁢     θ   0       +         P   gl     ⁢   λ       D   x           ⁢     
     ⁢         P   Gl     =     ±   1       ,       2   ⁢   …   ⁢           ⁢   with   ⁢           ⁢          sin   ⁢           ⁢     θ   gl              ≤   1     ,             (   6   )               
where θ gl , is the angle that the grating lobes appear and D x  is the element spacing.
 
Modeling
 
     Mutual coupling effects to a first order approximation can be described in terms of an active reflection coefficient which effects are shown in  FIG. 2  and written as 
     
       
         
           
             
               
                 
                   
                     Γ 
                     1 
                   
                   = 
                   
                     
                       
                         V 
                         refl 
                       
                       
                         V 
                         trans 
                       
                     
                     = 
                     
                       
                         
                           
                             
                               C 
                               1 
                             
                             ⁢ 
                             
                               S 
                               11 
                             
                           
                           + 
                           
                             
                               C 
                               2 
                             
                             ⁢ 
                             
                               S 
                               12 
                             
                           
                         
                         
                           C 
                           1 
                         
                       
                       = 
                       
                         
                           S 
                           11 
                         
                         + 
                         
                           
                             
                               C 
                               2 
                             
                             
                               C 
                               1 
                             
                           
                           ⁢ 
                           
                             S 
                             12 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   7 
                   ) 
                 
               
             
             
               
                 
                   
                     Γ 
                     2 
                   
                   = 
                   
                     
                       
                         V 
                         refl 
                       
                       
                         V 
                         trans 
                       
                     
                     = 
                     
                       
                         
                           
                             
                               C 
                               2 
                             
                             ⁢ 
                             S 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             22 
                           
                           + 
                           
                             
                               C 
                               1 
                             
                             ⁢ 
                             
                               S 
                               21 
                             
                           
                         
                         
                           C 
                           2 
                         
                       
                       = 
                       
                         
                           S 
                           22 
                         
                         + 
                         
                           
                             
                               C 
                               1 
                             
                             
                               C 
                               2 
                             
                           
                           ⁢ 
                           
                             S 
                             21 
                           
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   8 
                   ) 
                 
               
             
           
         
       
     
     The field pattern can then be described in terms of forward and backward traveling waves 
                       E   ϕ     ⁢   oc   ⁢           ⁢       E   element     ⁡     [             (       C   1     +       C   1     ⁢     S   12         )     ⁢     ⅇ       j   ⁡     (     kd   /   2     )       ⁢   sin   ⁢           ⁢   ϕ                     +     (       C   2     +       C   2     ⁢     S   22       +       C   1     ⁢     S   21         )       ⁢     ⅇ       j   ⁡     (       -   kd     /   2     )       ⁢   sin   ⁢           ⁢   ϕ               ]         ,           (   9   )               
where the excitation can be described in terms of the phase and voltage at the input terminals of the antenna written as
 
 C   1   =V   1   e   jφ     1      C   2   =V   2   e   kφ     2   ,  (10)
 
       FIG. 2  shows there is a difference between the experimental and the theoretical patterns. The most noticeable differences in the field patterns can be seen in lower levels of the field pattern or null locations. Improvements are found when the active impedance is taken into account. This can be attributed to the domination of the coupling parameters at null locations.  FIG. 3  shows the results for scanning at two different angles, including mutual coupling effects. 
     Hardware Design 
     A single element hardware setup for a variable phase shifter can be seen in  FIG. 4 . In  FIG. 4 , the variable phase shifter  10  includes a serial in, serial out shift register  16  which receives as input a reference clock  12  and feedback signal  14 , V PLL (t). A 16 to 1 multiplexer  18  is electrically connected to the shift register  16 . A register  20  is electrically connected to the multiplexer  18 . A shifted output signal, V SHIFTED (t) is provided into the phase locked loop  24 . The output, V PLL (t) is provided to an amplifier  26  which is electrically connected to an antenna element  28 . 
     The reference clock is divided down by 16 to provide a data source and is represented as 
                         V   ref     ⁡     (   t   )       =       π   4     ⁢       ∑       l   =   1     ,   3   ,   5   ,   …     oo     ⁢           ⁢       1   l     ⁢     sin   ⁡     (       l   ⁢           ⁢   π   ⁢           ⁢   t     L     )               ,           (   11   )               
where L is half the time period. The shift registers are shifted at the clock rate. The shift register contains 16 different delayed versions, sampled on the rising edge of the clock, as shown in Eq. 12.
 
                       V   shifted     ⁡     (     t   ,   i     )       =       π   4     ⁢       ∑       l   =   1     ,   3   ,   5   ,   …     oo     ⁢           ⁢       1   l     ⁢     sin   ⁡     (       1   ⁢     ω   0     ⁢   t     +       2   ⁢   π   ⁢           ⁢   i     16       )                     (   12   )               
for i=1, 2, . . . , 16 [6]. The pll (phase locked loop) locks into phase with the shifted data and provides a 2.425 GHz source and is represented as
 
 V   pll,p ( t,i )= B   p  sin(ω rf   t+φ   0   p ),  (13)
 
where,
 
                     ϕ   0   p     =         n   m     ⁢       2   ⁢   π   ⁢           ⁢   i     16       +     ϕ   loopdelay               (   14   )               
for i=1, 2, . . . , 16 where m and n are the frequency divide ratios of the reference and RF signal of the phase locked loop respectively as which is shown in  FIG. 5A . As shown in  FIG. 5A , a reference signal V REF  is provided to a divide by M counter  34  which is electrically connected to a phase detector  36 , the output of which is electrically connected to a filter  38 . The filter  38  is electrically connected to a voltage controlled oscillator (VCO)  40  which provides an output signal, V RF . Feedback is provided from the VCO  40 , through the divide by N counter  44  back to the phase detector  36 .  FIG. 5B  illustrates an alternative phase locked loop which introduces a phase aid into the phase locked loop to reduce the transient time needed for convergence by introducing a transient which, when in combination with the original response, produces a pseudo convergence of the loop. In  FIG. 5B , a phase detector  36  is electrically connected to a filter  38 . The output from the filter  18  is combined with a phase aid  39  to provide an input to the VCO  40 . Feedback from the VCO  40  is used as input to a programmable counter  45  which is electrically connected to an input of the phase detector  36 .
 
     The delayed versions of the baseband signal and the RF signal can be seen in  FIG. 6 . The division ratio produces a scaled version of the phase offset. The accuracy of Eq. 14 can be seen in  FIG. 7  which shows reasonable agreement with experimental results. The anomaly at integer 8 can be explained by a cycle slip of the registers. Scaling can be minimized or eliminated by using a frequency offset phase locked loop. In order to help ensure stability and zero steady state phase error during phase hops, a loop filter resulting in a third order loop was chosen [7]. However, other loop configurations could also be used for which these conditions are governing considerations. The settling time of the phase locked loop can be seen in  FIG. 8 . The signal is sent to a power amplifier whose desired load impedance is matched to the inactive input impedance of the antenna terminals. The input signal can be represented as
 
 V   antenna,p ( t,i )= C   p  sin(ω rf   t+φ   o   p )  (15)
 
     It will be shown that the phased array pattern is independent of a given modulation scheme. For example, a QPSK modulation scheme can be described in terms of the following excitation per symbol. 
                       A   _     baseband   p     =       A   p     ⁢     sin   ⁡     [         w   baseband     ⁢   t     +       (     i   -   1     )     ⁢     π   2       +     ϕ   0   p       ]                 (   16   )               
for i=1, 2, 3, 4 and the excitation coefficients at the antenna terminals can be represented as
 
     
       
         
           
             
               
                 
                   
                     
                       C 
                       _ 
                     
                     RF 
                     P 
                   
                   = 
                   
                     
                       C 
                       mod 
                     
                     ⁢ 
                     
                       C 
                       p 
                     
                     ⁢ 
                     
                       ⅇ 
                       
                         j 
                         ⁡ 
                         
                           [ 
                           
                             
                               
                                 ( 
                                 
                                   i 
                                   - 
                                   1 
                                 
                                 ) 
                               
                               ⁢ 
                               
                                 π 
                                 2 
                               
                             
                             + 
                             
                               ϕ 
                               0 
                               p 
                             
                           
                           ] 
                         
                       
                     
                   
                 
               
               
                 
                   ( 
                   17 
                   ) 
                 
               
             
           
         
       
     
     Inserting the excitation into Eq. 9, and after factoring, the field pattern can be written as 
                       E   ϕ     =       {             E   element     ⁡     [       (       C   1     +     C   11     +       C   2     ⁢     S   12         )     ⁢     ⅇ       j   ⁡     (     kd   /   2     )       ⁢   sin   ⁢           ⁢   ϕ         ]                   +     (       C   2     +       C   2     ⁢     S   22       +       C   1     ⁢     S   21         )       ⁢     ⅇ       j   ⁡     (       -   kd     /   2     )       ⁢   sin   ⁢           ⁢   ϕ               }     ⁢     C   mod     ⁢     ⅇ     j   ⁡     [       (     ⅈ   -   1     )     ⁢     k   2       ]             ,           (   18   )               
which is independent of the modulation angle. This can be generalized to any modulation scheme. The architecture presented is best suitable for QAM and QPSK modulations which are shown in  FIG. 9 . This architecture is suitable for high data rate transmission due to its ability to support QAM and QPSK modulation types.
 
Test Setup
 
       FIG. 10 , shows an automated setup  50  for power versus angle measurements. The spectrum analyzer  78  is connected to a computer  76 , which synchronizes the machine to an angular rotary device. As shown in  FIG. 10 , a V SHIFTED (θ 1 ) signal  52  is input into a first PLL  54 . The PLL  54  is electrically connected to a low pass filter  56 , which is electrically connected to an amplifier  58  which is electrically connected to an antenna element  60  which transmits a radio frequency signal  62 . Similarly, a V SHIFTED (θ 2 ) signal  64  is input into a second PLL  66  which is electrically connected to a low pass filter  68  which is electrically connected to an amplifier  70  which is electrically connected to another antenna element  72  which transmits a radio frequency signal  74  to a receive antenna  75  which is electrically connected to a bandpass filter  79  which is electrically connected to a spectrum analyzer  78  connected to the computer  76 . 
     The spectrum analyzer is configured for narrow band measurements that are averaged to reduce measurement variation by the square root of the average factor. The reduction in variation allows for low side lobe measurements to be performed. The exact phase differences between the input signals were measured using an oscilloscope and these signals can be described by the equation below
 
 V   1 ( t )= C   1  sin(ω rf   t+φ   1 ),  (19)
 
and,
 
 V   2 ( t )= C   2  sin(ω rf   t+φ   2 )  (20)
 
     Using Eq. 19 and 20 and scattering parameter measurement results of the amplifier, filter, and interconnecting cables one obtains
 
 V   p,antenna ( t )= C   p,antenna  sin(ω rf   t+φ   p,antenna )  (21)
 
where
 
 C   p,antenna   =C   p,filter   C   p,amp   C   p,cable   B   p   (22)
 
and
 
φ p,antenna =φ p,cable +φ p,amp +φ 0   p   (23)
 
     The scattering parameters of the array are directly measured and combined with Eq. 9 to predict field pattern measurements. 
       FIG. 11  provides a simplified block diagram of the present invention. A system  80  is shown which includes a phased array antenna  88  which is electrically connected to an upconverter circuit  86  which is electrically connected to a phase adjusting circuit  84 . In operation, the baseband signal  82  is phase shifted by the phase adjusting circuit  84 . The resulting signals are then upconverted with the upconverter circuit  86  and communicated to the phased array antenna  88 . 
     The phase array system disclosed describes a transmitting system but a receiving system or a transmitting/receiving system of similar architectures can be readily assembled by those of ordinary skill in the art using the same techniques for steering the array. In a receiving system, the upconverter, for example  86  of  FIG. 11  would become a down-converter, the output amplifiers, for example  58  and  70  of  FIG. 10 , would become low noise amplifiers with gain in the receive direction, and the phase adjusting circuit, for example,  84  of  FIG. 11  would become a phase detecting circuit. Those of ordinary skill in the art would know that for a transmitting/receiving system a diplexer/duplexer could become redundant but in general a diplexer/duplexer would be used in a full transmitting/receiving system. 
       FIG. 12  provides a simplified block diagram of the present invention for a receiving system. A system  90  is shown which includes a phased array antenna  88  which is electrically connected to a downconverter circuit  92  which is electrically connected to a phase detecting circuit  94 . In operation, signals are communicated from the phased array antenna  88  to the downconverter circuit  92 . Phase detection is performed by the phase detecting circuit  94 . 
       FIG. 13  provides a simplified block diagram of a transmitting/receiving system  102 . In  FIG. 13 , a diplexer  96  is electrically connected to the phased array antenna  88 . The diplexer  96  directs the transmitted signal from the transmit path and the received signal to the receive path. In the transmitting path, the baseband signal  82  is provided to the phase adjusting circuit  84  which is electrically connected to the upconverter circuit  86 . An output amplifier  98  is shown which is electrically connected to the diplexer  96 . On the receive side, the diplexer  96  is electrically connected to amplifier  100  which is a low noise amplifier. The amplifier  100  is electrically connected to the downcoverter circuit  92  which is electrically connected to the phase detecting circuit  94 . 
     Therefore, a method and system for a phased array antenna system has been disclosed, modeling methods to accurately predict beam formation have been described and a 2.425 GHz phased array architecture for automatic beam steering has been shown as well as suitable modulation techniques and an automated test setup with experimental techniques. The present invention contemplates numerous variations in the specific frequencies used, although of particular interest is frequencies above 1 GHz and preferably above 2 GHz; the type of antennas used for transmitting and receiving; the type of modulation used; and other variations, options, and alternatives. 
     It is also apparent to those of ordinary skill in the art that phase shift at a frequency is related to time delay of a signal as: 
               time   ⁢           ⁢   delay     =       1   360     ⁢         phase   ⁢           ⁢   delay     -     deg   ⁢           ⁢   rees         frequency   -   hertz               
such that when this disclosure speaks of phase shift or phase delay it could also speak of time shift or time delay.
 
     It is to be understood that the embodiments described herein are merely illustrative of the many possible specific arrangements that can be devised in application of the principles of the invention. Other arrangements can be devised in accordance with these principles by those of ordinary skill in the art without departing from the scope and spirit of the invention. It is therefore intended that such other arrangements be included within the scope of the following claims and their equivalents. 
     REFERENCES 
     All of the references cited in herein are hereby incorporated by reference in their entireties.
     [1] Balanis c., “Antenna Theory, Analysis and Design,” Wiley Interscience, pp. 816-843, 2005.   [2] Molisch, Andreas F., “Wireless Communications,” John Wiley and sons, pp. 80, July 2006.   [3] Weisbcck, Ing., “Lecture notes to Introduction to Microstrip Antennas,” University Karlsruhe pp. 58, 2001.   [4] D. M. Pozar, “The Active Element Pattern,”  IEEE Transactions on Antennas and Propagation , vol. 42, no. 8, August 1994.   [5] Wanner, Shannon, Weber, Robert 1., Song, Jiming, “Mutual Coupling in Phase Array”, Antennas and Propagation-Society, 2007   [6] Egen, William, “Phase Locked Basics,” Wiley Interscience, pp. 249, 1998.   [7] Donald R. Stephens, Phase-Locked Loops for Wireless Communications: Digital, Analog and Optical Implementations, Kluwer Academic Publishers, 2nd edition, 2001.