Abstract:
Methods and apparatus to facilitate improve code division multiple access (CDMA) receivers are disclosed. An example method disclosed herein comprises: receiving a signal containing first portions that are based on known data and second portions that are based on unknown data; generating a training signal, from the received signal, that substantially represents one or more of the first portions; adapting filter coefficients using the training signal; and equalizing the received signal using the adapted filter coefficients.

Description:
RELATED APPLICATION 
     This patent claims benefit of U.S. Provisional Application Ser. No. 60/601,375, entitled “Flexible Advanced Receiver for CDMA Systems” and filed on Aug. 12, 2004. U.S. Provisional Application Ser. No. 60/601,375 is hereby incorporated by reference in its entirety. 
    
    
     FIELD OF THE DISCLOSURE 
     This disclosure relates generally to receivers for wireless communication systems, and, more particularly, to methods and apparatus to facilitate improved code division multiple access (CDMA) receivers. 
     BACKGROUND 
     Third generation (3G) mobile radio standards (e.g., CDMA2000, Universal Mobile Telecommunications Systems (UMTS)) for wireless communication systems are the result of a massive worldwide effort involving many companies since the mid-1990s. 3G standards initially supported data rates up to 2 megabits per second (Mbps) but have evolved to support data rates up to 14 Mbps. 
       FIG. 1  is a diagram of an example prior-art wireless communication system  100  consisting of a wireless base station  110  and a plurality of wireless mobile devices (e.g., a wireless handset  115 , a laptop computer  116 , a personal digital assistant (PDA)  117 , etc.) with which the wireless base station  110  is capable of communicating data and/or voice information. 
       FIG. 2  is a block diagram illustration of an example prior-art manner of implementing the wireless handset  115  of  FIG. 1 . The wireless handset  115  contains an antenna  210  used to transmit and receive wireless radio frequency (RF) signals to and from the wireless base station  110  (not shown). To process the RF signals received from the wireless base station  110  via the antenna  210 , and to generate RF signals for transmission to the wireless base station  110  via the antenna  210 , the example wireless handset  115  contains a RF transceiver  215 . The RF transceiver  215  modulates baseband transmit signals received from the analog baseband processor  220  to RF band signals, and demodulates RF band signals received from the RF transceiver  215  to baseband. 
     To handle conversion from the analog domain to the digital domain, the example wireless handset  115  further includes an analog baseband processor  220 . The analog baseband processor  220  comprises an analog-to-digital (A/D) converter (not shown) to transform analog baseband signals received from the RF transceiver  215  into digital baseband signals for the digital baseband processor  225 . The analog baseband processor  220  also includes a digital-to-analog (D/A) converter (not shown) to transform digital baseband signals received from the digital baseband processor  225  into analog baseband signals for the RF transceiver  215 . 
     To implement the digital receive functions (e.g., equalization, despreading, demodulation, etc.) and the digital transmit functions (e.g., modulation, spreading, etc.) the example wireless handset  115  includes the digital baseband processor  225 . To encode and decode signals representative of speech, the example wireless handset  1   5  further includes a voice-coder-decoder (vocoder)  230 . The vocoder  230  comprises a speech encoder (not shown) to translate digital samples representing speech spoken by the user (not shown) of the wireless handset  115  into a stream of digital data to be processed for transmission to the wireless base station  110  by the digital baseband processor  225 , the analog baseband processor  220 , the RF transceiver  215 , and the antenna  210 . Likewise, the vocoder  230  comprises a speech decoder (not shown) to translate a stream of digital data received from the wireless base station  110  into digital samples representative of speech to be listened to by the user of the wireless handset  115 . 
     The example wireless handset  115  further comprises a voice transceiver  235  that implements conversion of analog signals representative of speech received from a microphone  245  into digital samples using, among other things, an A/D converter (not shown). The voice transceiver  235  further implements conversion of digital samples representative of speech received from the vocoder  230  to analog signals which can be played out a speaker  240  using, among other things, a D/A converter (not shown). 
     Example implementations of the antenna  210 , the RF transceiver  215 , the analog baseband processor  220 , the vocoder  230 , the voice transceiver  235 , the speaker  240 , and the microphone  245  are well known to persons of ordinary skill in the art, and, thus, will not be discussed further. 
     Asymmetric user services (e.g., web browsing) requiring high downlink capacity (i.e., from wireless base station to the wireless mobile device) led to the development of the High Speed Downlink Packet Access (HSDPA) and the Evolution Data and Voice (EV-DV)/Evolution Data Optimized (EV-DO) standards. Efficient downlink wireless receivers that can operate in the presence of multiple transmission paths are important to achieving high downlink capacity. In subsequent discussions the term multipaths refers collectively to a plurality of transmission paths by which a signal transmitted by one device is received by a second device. In such circumstances, a receiver receives the transmitted signal a number of times wherein each received version of the transmitted signal has a different time delay, signal attenuation, and phase relative to the other received versions. Multipaths are created by reflections of the wireless signal off of objects located near the transmitter, the receiver, or in between. For example, buildings, bridges, cars, clouds, etc. Further, the term multipath refers to one of the plurality of transmission paths between the first device and the second device (i.e., one of the paths comprising the multipaths). 
     It is widely recognized that the well-known, conventional RAKE receiver suffers significant degradation in performance in the presence of multipaths on the downlink of a wireless communication system. In 3G standards, code transmissions on the downlink are typically designed to be orthogonal at a transmitter, but multipaths destroy the orthogonality resulting in significant inter-code interference at a wireless receiver. In the context of 3G, orthogonality refers to the use of spreading sequences that are orthogonal to each other, i.e., there is no correlation between a first spreading sequence and a second spreading sequence. The lower the CDMA spreading factor being utilized and the closer a user is located to the wireless base station  110 , the more deleterious the effect of interference due to multipaths becomes. High speed data wireless systems such as EV-DV/EV-DO and HSDPA for UMTS employ a low spreading factor and may utilize a scheduling rule that tends to select users that are close to the wireless base station  110 . The RAKE receiver does little to mitigate the effects of the multipath interference, therefore, the effects of multipath interference on these wireless mobile devices is quite significant. 
     Wireless receivers that try to mitigate the multipath interference in some fashion are termed advanced receivers. Two broad classes of CDMA advanced wireless receivers are equalizers and interference cancellers. If a received chip level signal, which has been distorted by multipaths, is sufficiently equalized prior to correlation with a spreading code, there is only a single path present during a subsequent despreading operation. A chip represents a time duration of a CDMA signal corresponding to one value in a pseudo-noise (PN) code sequence used to spread/de-spread the CDMA signal. Assuming orthogonal spreading sequences, an equalizer can largely restore orthogonality of multiple codes that started out orthogonal at the transmitter even in the presence of multiple transmission paths. Implementing equalization of chip level signals, rather than symbol level signals, allows the application of equalization to wireless communication systems utilizing long spreading sequences. 
     A downlink of a CDMA system typically uses a broadcast common pilot channel (CPICH) to transmit CPICH symbols that are known a priori to the wireless mobile device. The CPICH is used by a receiver in a wireless mobile device for synchronization, channel estimation, handoff support, etc. Additionally, in the case of an equalizer, the CPICH symbols may be used to train an equalizer. Frequently the normalized least mean squares (NLMS) algorithm is used to adapt equalizer coefficients. For simplicity, we shall refer to such CPICH trained NLMS equalizers as “CPICH based NLMS”. 
       FIG. 3  is a block diagram illustration of an example prior-art manner of implementing the digital baseband processor  225  of  FIG. 2 . The digital baseband processor  225  generally includes a transmitter  310  and a receiver  315 . To convert digital data representative of encoded speech received from the vocoder  230  into CDMA transmit signals, the transmitter  310  includes a channel encoder  320 , a modulator  322 , and a spreader  325 . The channel encoder  320  includes, among other things, the following functions: channel encoding, cyclic redundancy check (CRC) generation, conversion to blocks of data, rate matching, interleaving, multiplexing, etc. The modulator  322  modulates (e.g., using QPSK, QAM, etc.) the output of the channel encoder  320  which is then multiplied with a PN code sequence by the spreader  325  to create a digital CDMA transmit signal. The PN code sequence is formed as an exclusive-or of a cell specific PN code sequence and a channel specific PN code sequence. 
     To help mitigate the effects of multipath interference, the example receiver  315  includes an equalizer  328  to apply a filter to an input chip level signal  326 . As discussed above, the equalizer  328  works to restore orthogonality of signals being received by the receiver  315 . Filter coefficients of the equalizer  328  are adapted using an error signal  331  formed as a difference between a spread pilot signal  332  and an output  329  of the equalizer. The spread pilot signal  332  is formed by multiplication of CPICH symbols and a first PN code sequence. The CPICH symbols are those transmitted by the wireless base station  10  on the CPICH. The first PN code sequence is an exclusive-or of the cell specific PN code sequence and a CPICH specific code sequence. Because the duration of each CPICH symbol is multiple (e.g., 256) chips, the multiplication of the CPICH symbols and the first PN code sequence multiplies each CPICH symbol by N chips of the first PN code sequence, where N is the duration of each CPICH symbol. 
     If the equalizer  328  adapts its filter coefficients using NLMS, then the equalizer update configuration shown in  FIG. 3  (comprising the equalizer  328 , the equalizer output  329 , the spread pilot signal  332 , and the error  331 ) is the conventional, prior-art “CPICH based NLMS.” In a conventional “CPICH based NLMS” chip level equalizer, the input chip level signal  326  used in equalizer training comprises not only multipath signals of the CPICH, but also signals from one or more other downlink channels. At a chip level these signals are all roughly the same power, thus, the variance of the error signal  331  used in equalizer training will be quite large. Such a large variance limits the achievable performance of the equalizer  328 . 
     Because each CDMA transmit signal has been spread by multiplication with a second PN code sequence (formed as an exclusive-or of the cell specific PN code sequence and the data channel specific code sequence) in the wireless base station  110 , the receiver  315  includes a despreader  330  that correlates the equalized received signal  329  with the second PN code sequence. This second PN code sequence is the same PN code sequence used in the wireless base station  10  to spread data symbols currently being received. Outputs of the correlation process are provided every spreading factor (SF) chips. If adequate equalization is not performed in the presence of multipaths, every significant multipath must be individually despread by the despreader  330 . 
     The outputs of the despreader  330  are passed through a demodulator  335  before being passed to a channel decoder  340  that performs, among other things: de-multiplexing, de-interleaving, rate detection and de-rate matching, conversion from blocks, CRC checking, channel decoding, etc. 
     Example implementations of the channel encoder  320 , the modulator  322 , the spreader  325 , the equalizer  328 , the despreader  330 , the demodulator  335 , and the channel decoder  340  are well known to persons of ordinary skill in the art, and, thus, will not be discussed further. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a diagram of an example prior-art wireless communication system with a wireless base station and a plurality of wireless mobile devices. 
         FIG. 2  is a block diagram illustration of an example prior-art manner of implementing the wireless handset of  FIG. 1 . 
         FIG. 3  is a block diagram illustration of an example prior-art manner of implementing the digital baseband processor of  FIG. 2 . 
         FIG. 4  is a block diagram illustration of a portion of a disclosed example digital baseband receiver. 
         FIG. 5  is a graph illustrating data throughput improvements of the example digital baseband receiver of  FIG. 4 . 
         FIG. 6  is a block diagram illustration of a disclosed example manner of implementing the channel estimator of  FIG. 4 . 
         FIG. 7  is a block diagram illustration of a disclosed example manner of implementing the pilot regenerator of  FIG. 4 . 
         FIG. 8  is a block diagram illustration of a disclosed example manner of implementing the equalizer trainer of  FIG. 4 . 
         FIG. 9  is a block diagram illustration of a portion of a disclosed example digital baseband receiver, in which a wireless base station implements space time transmit diversity (STTD). 
         FIG. 10  is a block diagram illustration of a disclosed example manner of implementing the TX Antenna 2  Adjuster of  FIG. 9 . 
         FIG. 11  is a block diagram illustration of a portion of a disclosed example digital baseband receiver, in which a wireless base station implements closed loop transmit diversity (CLTD). 
         FIG. 12  is a block diagram illustration of a portion of a disclosed example digital baseband receiver using receiver diversity. 
         FIG. 13  is a block diagram illustration of a disclosed example manner of implementing the equalizer trainers of  FIG. 12 . 
         FIG. 14  is a block diagram illustration of a portion of a disclosed example digital baseband receiver, in which the receiver is configurable to implement improved equalizer training, conventional equalizer training, or a RAKE receiver. 
         FIGS. 15   a - c  illustrate the relationship between channel estimates and finite impulse response (FIR) filter coefficients in a RAKE receiver. 
         FIG. 16  a flow chart illustrating a disclosed example process that may be executed by the digital baseband receiver of  FIG. 4 . 
         FIG. 17  is a block diagram illustration of a disclosed example digital signal processor that may execute the process of  FIG. 16  to implement the digital baseband receiver of  FIG. 4 . 
     
    
    
     DETAILED DESCRIPTION 
     A conventional CPICH based chip-level NLMS equalizer directly applies time domain techniques from narrow band systems (e.g., GSM) that, while often better than a RAKE receiver for low data rates, are less effective in high data rate CDMA systems since, by design, CDMA signals have low signal to noise ratio (SNR) characteristics at the chip level.  FIG. 4  is a block diagram illustrating a portion of a disclosed example digital baseband receiver  400 . In a CDMA system, the input chip level signal  326  is comprised of signals associated with a number of downlink channels (interferers), multipaths, and additive channel noise. Taking advantage of the interference suppression capabilities of CDMA spreading sequences, the illustrated example includes a multipath pilot signal generator  401  to generate an equalizer training signal  430  (from the input chip level signal  326 ) substantially representative of CPICH symbols that have been spread and then received through multiple transmission paths (i.e., a multipath pilot signal). Interferers (e.g., other control and data channels) and additive channel noise present in the training signal  430  are substantially reduced compared to the input chip level signal  326 . Thus, the training signal  430  has a substantially improved SNR compared to the input chip level signal  326  such that, an equalizer whose coefficients are adapted using the training signal  430  performs substantially better than a conventional prior-art CPICH based NLMS equalizer. 
     The equalizer training signal  430  is generated by the multipath pilot signal generator  401  by (a) determining channel estimates  412  from the input chip level signal  326 , (b) re-spreading the channel estimates  412  to form spread channel estimates  422 , and (c) combining the spread channel estimates  422  using a summer  425 . To determine the channel estimates  412 , the multipath pilot signal generator  401  includes a channel estimator  405  which despreads (using a PN code sequence  408  generated by a PN generator  410 ), accumulates, and filters the input chip level signal  326  to extract received pilot channel symbols (for each multipath). As will be discussed below in conjunction with  FIG. 6 , the despreading process implemented in the channel estimator  405  accounts for the timings of each multipath. Further, long enough accumulations are used to properly balance multipath interference against channel tracking performance. 
     The PN generator  410  forms the PN code sequence  408  as an exclusive-or of the CPICH specific code sequence and the cell specific PN code sequence. For UMTS the CPICH specific code sequence is the all zeros sequence (for the primary CPICH), and, thus, can be ignored and the exclusive-or can be eliminated. 
     To spread the channel estimates  412 , the multipath pilot signal generator  401  provides a pilot regenerator  415  to multiply (using correct timings for each multipath) each of the channel estimates  412  by a delayed PN code sequence  417 . The receiver  400  provides a delay  420  to delay the PN code sequence  408  by an amount corresponding to filtering delays in the channel estimator  405  to create the delayed PN code sequence  417 . The multipath pilot signal generator  401  further provides a summer  425  to add together each of the re-spread channel estimates  422  to form the equalizer training signal  430 . 
     To determine equalizer coefficients (to restore orthogonality of received signals), the example digital baseband receiver  400  of  FIG. 4  includes an equalizer trainer  435 . The equalizer trainer  435  applies a filter (whose coefficients are substantially equivalent to coefficients  455  of an equalizer  445 ) to the training signal  430  (i.e., the multipath pilot signal) to form an equalized multipath pilot signal. The equalizer trainer  435  adapts the coefficients of the filter to minimize an error between the equalized multipath pilot signal and a locally generated pilot training signal  437 . The locally generated pilot signal  437  is formed by multiplying CPICH symbols  436  by a further delayed PN code sequence  442 . The further delayed PN code sequence  442  is formed by further delaying the delayed PN code sequence  417 , using a delay  440 , where the amount of delay implemented by the delay  440  corresponds to the centering delay of the multipath channels. Since the duration of each CPICH symbol  436  is multiple (e.g., 256) chips, the multiplication of CPICH symbols  436  and the further delayed PN code sequence  442  multiplies each CPICH symbol  436  by N chips of the further delayed PN code sequence  442 , where N is the duration of each CPICH symbol  436 . For UMTS the CPICH symbols  436  are all identical, however, because the CPICH symbol is complex-valued the multiplication can not be eliminated. 
     Under the assumptions that CPICH symbols  436  are identical, the training signal  430  substantially comprises a summation of the channel estimate  412  for each multipath spread by the PN code sequence  408 ; and scaled and rotated by the CPICH symbol  436 . Further, the locally generated pilot signal  437  comprises the PN code sequence  408  scaled and rotated by the CPICH symbol  436 . As constructed, the training signal  430  and the locally generated pilot signal  437  substantially only differ in whether or not they include the channel estimates  412 . Thus, the equalizer trainer  435  properly adapts coefficients to equalize the multipath channel responses. If, the CPICH symbols  436  are not identical, then the training signal  430  substantially comprises a summation of CPICH symbols  436  convolved with the channel estimate  412  for each multipath further spread by the PN code sequence  408 . Because the locally generated pilot signal  437  includes the multiplication with CPICH symbols  436 , the equalizer trainer  435  will properly adapt coefficients to equalize the multipath channel responses. 
     In the illustrated example, the filter of the equalizer trainer  435  is a finite impulse response (FIR) filter. Further, the adaptation of the coefficients uses NLMS. Filters (e.g., FIR) and filter coefficient adaptation (e.g., NLMS) are well known to persons of ordinary skill in the art, and, in the interest of brevity, are not discussed further. Persons of ordinary skill in the art will readily appreciate that the filter could be implemented using some other suitable filter structure or filtering arrangement; and/or other forms of coefficient adaptation (e.g., least mean squares (LMS), recursive least squares (RLS), etc.) could be implemented. 
     To receive the user data present in the input chip level signal  326 , the digital baseband receiver  400  further comprises a delay  450  and a despreader  460 . The delay  450  delays the input chip level signal  326  by an amount corresponding to any filtering delays present in the channel estimator  405  creating a delayed input chip level signal  452 . To equalize the delayed input chip level signal  452  the equalizer  445  applies a filter to the delayed input chip level signal  452  creating an equalized received signal  454 . In the illustrated example, the filter of the equalizer  445  is an FIR filter. However, persons of ordinary skill in the art will readily appreciate that the filter could utilize other suitable filter structures or filtering arrangements. In the illustrated example the structure of the filter of the equalizer trainer  435  is the same as the structure of the filter of the equalizer  445 ; and the coefficients  455  of the filter of the equalizer  445  are substantially equal to the coefficients of the filter of the equalizer trainer  435 . However, it should be appreciated by persons of ordinary skill in the art that the two filters could utilize different filter structures and filter coefficients as long as they realize substantially equivalent transfer functions. 
     To despread the equalized received data signal  454 , the despreader  460  multiplies the equalized received signal  454  with a second PN code sequence that is an exclusive-or of the cell specific PN code sequence and a data channel specific code sequence. Finally, further receive processing (not shown) of the output  462  of the despreader  460  is performed. For example, in the case of HSDPA channels, the output  462  of the despreader  460  is processed to extract symbols corresponding to the HSDPA channels and is followed by the calculation of log likelihood ratios that characterize the reliability of those symbols. Typically, hybrid-ARQ combining and turbo decoding are performed next to recover transmitted information bits corresponding to user data. 
     In the illustrated example just discussed, a precise multipath profile (e.g., multipath timings, signal strengths, etc.) is utilized. Alternatively, to despread the input chip level signal  326  several contiguous chip spaced paths (regardless of whether any signal energy exists on some of them or not) may be used. 
       FIG. 5  shows a graph  500  showing exemplary data throughput improvements in Kbps resulting from the illustrated example digital baseband receiver  400  of  FIG. 4 .  FIG. 5  illustrates the performance, for a variety of channel signal to noise conditions (represented by IOR/IOC as recorded in decibels (dB)), of (a) the example digital baseband receiver  400  of  FIG. 4  using line  505 , (b) a conventional RAKE receiver using line  510 , and (c) a CPICH based NLMS receiver using line  515 , where IOR is the total received signal power from the serving wireless base station and IOC is the total inteference power from all other (i.e., non-serving) wireless base stations. The ratio IOR/IOC reflects the geometry of the signal and roughly equates to the SNR of the signal. For a low IOR/IOC ratio, the SNR of training signals is often better than the SNR of data carrying signals. This causes a condition where equalizer coefficients adapted during training may be sub-optimal for data reception. Thus, for low IOR/IOC ratio conditions a RAKE receiver may perform better than either the example digital baseband receiver  400  of  FIG. 4  or a CPICH based NLMS receiver. However, a simple adaptation loop that makes the adaptation step-size dependent on the SNR could be used. For example, at low SNR with the CPICH based NLMS or the example digital baseband receiver  400  of  FIG. 4 , a smaller adaptation step-size (compared to high SNR) could be used, thus, providing better noise suppression and improved performance at the expense of adaptation speed. An example multi-mode solution, as discussed below in connection with  FIG. 14 , allows the use of either a RAKE receiver or the example digital baseband receiver  400  of  FIG. 4 , thus, achieving the best performance regardless of SNR. 
       FIG. 6  is an illustration of a disclosed example manner of implementing the channel estimator  405  of  FIG. 4 . To despread the input chip level signal  326  and to eliminate interference from non-pilot channels (e.g., other, undesired, downlink channels) into each multipath and to suppress interference from other multipaths, the channel estimator  405  provides a plurality of despreaders  605 ,  606 , and  607 . A despreader  605 ,  606 , and  607  is provided for each of one or more multipaths present in the input chip level signal  326 . The despreaders  605 ,  606 , and  607  despread the input chip level signal  326  by multiplication of the input chip level signal  326  with a delayed PN code sequence specific to each multipath. To form the delayed cell specific PN code sequences for each multipath, the channel estimator  405  further provides a plurality of delay elements  610 ,  611 , and  612 . The delay elements  610 ,  611 , and  612  delay the PN code sequence  408  by an amount substantially corresponding to the multipath timing for each multipath. The detection of multipaths and the adaptation of their timings are well known to persons of ordinary skill in the art, and, thus, will not be discussed further. 
     To extract received symbols, the example channel estimator  405  of  FIG. 6  includes a plurality of chip accumulators  615 ,  616 , and  617 . The chip accumulators  615 ,  616 , and  617  perform an accumulation over a number (e.g.,  256 ) of chips to extract despread pilot channel symbols  650  (i.e., received symbols). The despread pilot channel symbols  650  for each multipath will substantially comprise transmitted CPICH symbols  436  convolved with a channel response for the multipath. If the transmitted CPICH symbols are identical, the despread pilot channel symbols  650  substantially comprise channel responses of each multipath scale and rotated by the CPICH symbol  436 . In the illustrated example, the summation is performed over  256  chips. 
     When the wireless base station  110  is using two-antenna transmit diversity in UMTS, then first CPICH symbols transmitted by a first transmit antenna are orthogonal to second CPICH symbols transmitted by a second antenna only when descrambled and summed over 512 chips. If transmit diversity is used, then in the illustrated example after despreading and summing over 256 chips accumulation of the despread pilot symbol signals  650  over two symbols is performed. Thus, effectively implementing despreading over 512 chips. If transmit diversity is not used by the wireless base station, then accumulation over 2 symbols need not be performed. To implement the accumulation over 2 symbols, the channel estimator  405  comprises a plurality of symbol summers  620 ,  621 , and  622 . In the illustrated example, the symbol summers  620 ,  621 , and  622  only operate when transmit diversity is used by the wireless base station  110 . Alternatively, for simplicity in the architecture, the symbol summers  620 ,  621 , and  622  are active regardless of whether or not the wireless base station is using transmit diversity. It should be readily apparent to persons of ordinary skill in the art that the number of chips accumulated by the chip accumulators  615 ,  616 , and  617  and the number of symbols summed by the symbols summers  620 ,  621 , and  622  can be modified from the illustrated example as long as the chosen values ensure orthogonality of the first and second CPICH symbols transmitted by the two transmit antennas. 
     To obtain reliable channel estimates  412  for each multipath the channel estimator  405  further comprises a plurality of filters  625 ,  626 , and  627 . In the illustrated example, the plurality of filters  625 ,  626 , and  627  are implemented as infinite impulse response (IIR) filters and provide a low-pass filtering (LPF) transfer function. The LPF transfer function serves to reduce out-of-band noise and to help smooth the channel estimates. It should be readily apparent to persons of ordinary skill in the art that the filters  625 ,  626 , and  627  could utilize FIR filters or other suitable filter structures or filtering arrangements. 
       FIG. 7  is an illustration of a disclosed example manner of implementing the pilot regenerator  415  of  FIG. 4 . So that the multipath pilot signal  430  can be generated (by the summer  425 ), the pilot regenerator  415  comprises a plurality of spreaders  705 ,  706 , and  707 . A spreader  705 ,  706 , and  707  is provided for each of one or more multipaths present in the input chip level signal  326 . The spreaders  705 ,  706 , and  707  spread each of the channel estimates  412  (as provided by the channel estimator  405 ) by multiplication of each channel estimate  412  with a delayed PN code sequence specific to each multipath. To form the delayed PN code sequences, the pilot regenerator  415  further provides a plurality of delay elements  710 ,  711  and  712 . The delay elements  710 ,  711  and  712  further delay the delayed PN code sequence  417  by an amount substantially corresponding to the multipath timing for each multipath. 
       FIG. 8  is an illustration of a disclosed example manner of implementing the equalizer trainer  435  of  FIG. 4 . To create a signal  802  which substantially matches the locally generated pilot signal  437 , the equalizer trainer  435  provides a filter  805 . To form an error signal  815  useful for adapting coefficients of the filter  805 , the equalizer trainer  435  further provides a summer  810 . The summer  810  computes the error signal  815  as a difference between (a) the locally generated pilot signal  437  and (b) the output  802  of the filter  805 . Further, the filter  805  is an FIR filter and adaptation of the coefficients of the filter  805  uses NLMS. However, persons of ordinary skill in the art will readily appreciate that the filter  805  could utilize other suitable filter structures or filtering arrangements; and/or other forms of coefficient adaptation (e.g., LMS, RLS, etc.) could be implemented. 
     In the illustrated example the structure of the filter  805  is the same as the structure of the filter of the equalizer  445 ; and the coefficients of the filter of the equalizer  445  are substantially equal to the coefficients of the filter  805 . However, it should be appreciated by persons of ordinary skill in the art that the two filters could utilize different filter structures and filter coefficients as long as they realize substantially equivalent transfer functions. 
     Persons of ordinary skill in the art will readily appreciate that any known downlink channel (i.e., a channel for which the wireless receiver  315  knows a priori the transmitted symbols) could be used instead of the CPICH. Further, if two or more known downlink channels (possibly including the CPICH) were available, training signal  430  and signal  437  could be generated to represent the known data from the known downlink channels. For example, if two known downlink channels using different channel PN code sequences are used, the channel estimator  405  is expanded to include a second set of despreaders, accumulators, symbol summers, and filters. The first set would generate first channel estimates for the first known channel using a first PN code sequence; the second set would generate second channel estimates for the second known channel using a second PN code sequence. Likewise, the pilot regenerator  415  is expanded to include a second set of spreaders. The first set would spread the first channel estimates; the second set would spread the second channel estimates. The summer  425  would sum together all the outputs of the first and second sets of spreaders. Finally, the signal  437  is generated by summing together a first output of a first multiplier multiplying the delayed first PN code sequence with symbols associated with the first known channel and a second output of a second multiplier multiplying the delayed second PN code sequence with symbols associated with the second known channel. It will be further appreciated by persons of ordinary skill in the art, that alternative or multiple known downlink channels can be also be readily utilized with transmit and receive diversity. 
       FIG. 9  is a block diagram illustrating a portion of a disclosed example digital baseband receiver  900  communicating with the wireless base station  110  implementing space time transmit diversity (STTD). To receive signals from a second transmit antenna (not shown) at the wireless base station  110 , the example digital baseband receiver  900  provides a second pilot regenerator  915  generating a second plurality of spread channel estimates  922 , a second summer  925  providing a second equalizer training signal  930 , a second equalizer trainer  935  adapting a second set of equalizer coefficients  955 , a second equalizer  945  applying a filter (whose coefficients are substantially equivalent to the second set of equalizer coefficients  955 ) to the delayed input chip level signal  452 , and a second despreader  960  despreading an output  954  of the second equalizer  945 . The signals received, processed and generated by these additional blocks are associated with signals and symbols transmitted by the second transmit antenna. The implementations of, and relationships between, the second pilot regenerator  915 , the second summer  925 , the second equalizer trainer  935 , the second equalizer  945 , and the second despreader  960  are as discussed above in relation to  FIGS. 4 , and  6 - 8 . 
     In the wireless base station, second symbols to be transmitted by the second antenna are formed by multiplying a TX antenna 2  symbol pattern  982  by first symbols to be transmitted by the first antenna. For example, if CPICH symbols  436  (i.e., first symbols) for the first transmit antenna are identical, then CPICH symbols  981  (i.e., second symbols) transmitted by the second transmit antenna are the TX antenna 2  symbol pattern  982  scaled and rotated by the CPICH symbol  436 . To extract the portion of the input chip level signal  326  associated with the second transmit antenna and to generate channel estimates  912  associated with the second transmit antenna, the receiver  900  provides a TX antenna 2  adjuster  905 . The TX antenna 2  adjuster  905  multiplies each of the despread pilot channel symbols  650  (provided by the channel estimator  405  as discussed above in relation to  FIG. 6 ) with the TX antenna 2  symbol pattern  982  (known a priori to the receiver  900 ). 
     To adapt the equalizer coefficients  955  based on the second CPICH symbols  981  received from the second transmit antenna, the further delayed PN code sequence  442  is modified so as to be representative of the second CPICH symbols  981 . To modify the further delayed PN code sequence  442 , the example receiver  900  provides a multiplier  980 . The multiplier  980  multiples the further delayed PN code sequence  442  with the second CPICH symbols  981  to generate a second locally generated pilot signal  937  used by the equalizer trainer  935  to adapt the second set of equalizer coefficients  955 . Since the duration of each second CPICH symbol  981  is multiple (e.g.,  256 ) chips, the multiplication of second CPICH symbol  981  and the further delayed PN code sequence  442  multiplies each second CPICH symbol  981  by N chips of the further delayed PN code sequence  442 , where N is the duration of each second CPICH symbol  981 . 
     The second training signal  930  substantially comprises a summation of the second CPICH symbols  981  convolved with the channel estimate  912  for each multipath further spread by the PN code sequence  408 . Further, the locally generated pilot signal  937  (which includes the multiplication with second CPICH symbols  981 ) comprises the second CPICH symbols  981  spread by the PN code sequence  408 . In this configuration, the equalizer trainer  935  will properly adapt coefficients  955  to equalize the multipath channel responses associated with the second transmit antenna. 
     To combine the equalized and despread signals  462  and  962  received from the two transmit antennas (i.e., the output of despreaders  460  and  960 ), the example receiver  900  provides a STTD decoder  990 . As is well known to persons of ordinary skill in the art, the STTD decoder  990  buffers two symbols at a time from each of the signals  462  and  962  and combines them to form two symbols  995  that are provided to remaining portions (not shown) of a wireless receiver for further processing. 
       FIG. 10  is an illustration of a disclosed example manner of implementing the TX antenna 2  adjuster  905  of  FIG. 9 . To generate channel estimates  912  associated with the second transmit antenna (not shown) of the wireless base station  110  (not shown), the TX antenna 2  adjuster  905  provides a plurality of multipliers  1005 ,  1006 , and  1007 . In the illustrated manner, the multipliers  1005 ,  1006 , and  1007  multiply each of despread pilot channel symbols  650  (provided by the channel estimator  405  as discussed above in relation to  FIGS. 6 and 9 ) with the TX antenna 2  symbol pattern  982 . 
     As discussed above, the first CPICH symbols  436  transmitted by the first transmit antenna are orthogonal to the second CPICH symbols  981  transmitted by the second transmit antenna only when despread and accumulated over  512  chips. Because the chip accumulators  615 ,  616 , and  617  of  FIG. 6  each accumulated  256  chips, the TX antenna 2  adjuster  905  provides a plurality of symbol summers  1010 ,  1011 , and  1012 . In the illustrated example, each symbol summer  1010 ,  1011 , and  1012  determines a sum over two symbols. Thus, effectively implementing despreading over  512  chips. It should be readily apparent to persons of ordinary skill in the art that the number of chips accumulated by the chip accumulators  615 ,  616 , and  617  and the number of symbols summed by the symbols summers  1010 ,  1011 , and  1012  can be modified from the illustrated example as long as the chosen values ensure orthogonality of the CPICH symbols  436  and  981  transmitted by both transmit antenna of the wireless base station. 
     To obtain reliable channel estimates  912  for each multipath associated with the second transmit antenna of the wireless base station, the TX antenna 2  adjuster  905  further provides a plurality of filters  1020 ,  1021 , and  1022 . In the illustrated example, the plurality of filters  1020 ,  1021 , and  1022  are implemented as IIR filters and provide a low-pass filtering transfer function. The low-pass filtering transfer function serves to reduce out-of-band noise and to help smooth the channel estimates. It should be readily apparent to persons of ordinary skill in the art that the filters  1020 ,  1021 , and  1022  could utilize FIR filters or other suitable filter structures or filtering arrangements. 
       FIG. 11  is a block diagram illustrating a portion of a disclosed example digital baseband receiver  1100  communicating with the wireless base station  110  implementing closed loop transmit diversity (CLTD). The implementation of the illustrated example of  FIG. 11  is substantially the same as the illustrated example of  FIG. 9 . The difference between the illustrated examples of  FIGS. 9 and 11  is that different receive processing is applied following the despreaders  460  and  960 . As is well known to persons of ordinary skill in the art, the receiver  1100  is provided with a summer  1190  to combine the two equalized and despread signals  462  and  962 . Further, to allow the example receiver  1100  to combine the two equalized and despread signals  462  and  962  in a substantially optimal fashion, the receiver  1100  provides a multiplier  1185  and a weight verification unit  1180 . The weight verification unit  1180  performs antenna verification to determine a substantially optimum weighting of the two equalized and despread signals  462  and  962 . To balance the contribution of each of the equalized and despread signals  462  and  962 , the multiplier  1185  multiplies the output  962  of the second despreader  960  by the output of the weight verification unit  1180 . The output of the multiplier  1185  is then summed with the output  462  of the first despreader  460  to determine a substantially optimal received signal  1195 . The substantially optimal received signal  1195  is provided to remaining portions (not shown) of a wireless receiver for further processing. 
       FIG. 12  is a block diagram illustrating a portion of a disclosed example digital baseband receiver  1200  using two antenna receive diversity. To process a second input chip level signal  1226  from a second receive antenna, the example digital baseband receiver  1200  provides a second channel estimator  1205  to generate a second plurality of channel estimates  914 , a second pilot regenerator  915  generating a second plurality of scrambled channel estimates  922 , a second summer  925  providing a second equalizer training signal  930 , a second delay  1250 , a second equalizer  945  applying a filter (utilizing a second set of equalizer coefficients  955 ) to a second delayed input chip level signal  1252 , and a second despreader  960  despreading the output  954  of the second equalizer  945 . The signals received, processed and generated by these additional blocks will be associated with the second input chip level signal  1226  received by the second receive antenna. The implementations of, and relationships between, the second channel estimator  1205 , the second pilot regenerator  915 , the second summer  925 , the second delay  1250 , the second equalizer  945 , and the second despreader  960  are as discussed above in relation to  FIGS. 4 , and  6 - 8 . 
     To adapt equalizer coefficients, the example receiver  1200  further includes equalizer trainers  1270  and  1275  adapting the first and the second sets of equalizer coefficients  455  and  955 . To adapt the first equalizer coefficients  455  the first equalizer trainer  1270  applies a filter (whose coefficients are substantially equivalent to the current first equalizer coefficients  455 ) to the training signal  430  (i.e., the multipath pilot signal determined based on the input chip level signal  326  associated with the first receive antenna) to form a first equalized multipath pilot signal. The equalizer trainer  1270  adapts the coefficients of the filter to minimize an error between (a) the locally generated pilot training signal  437  and (b) a sum  1282  of the first equalized multipath pilot signal and a second equalized multipath pilot signal (determined by the second equalizer trainer  1275 ). In a similar fashion the second equalizer  1270  adapts the second equalizer coefficients  955  based on the locally generated pilot training signal  437 . 
     In the illustrated example the filters of the equalizer trainers  1270  and  1275  are FIR filters. Further, the adaptation of the coefficients uses NLMS. Persons of ordinary skill in the art will readily appreciate that: the filters could be implementing using some other suitable filter structures or filtering arrangements; and/or other forms of coefficient adaptation (e.g., LMS, RLS, etc.) could be implemented. 
     To combine the equalized and despread signals  462  and  962  received from the two receive antennas, the example receiver  1200  provides a summer  1285 . As is well known to persons of ordinary skill in the art, the summer  1285  combines the equalized and despread signals  462  and  962  to form a signal  1295  provided to remaining portions (not shown) of a wireless receiver for further processing. 
       FIG. 13  is an illustration of a disclosed example manner of implementing the equalizer trainer  1270  of  FIG. 12 . To create the signal  1282  which substantially matches a locally generated pilot signal  437 , the equalizer trainer  1270  provides a filter  805 . To form an error signal  1315  useful for adapting the coefficients of the filter  805 , the equalizer trainer  1270  further provides a summer  1310 . In the illustrated example, the summer  1310  computes the error signal  1315  as a difference between (a) a locally generated pilot signal  437  and (b) the sum  1282  of an output  1237  of the filter  805  and a second signal  1286 . Further, the filter  805  is an FIR filter and adaptation of the coefficients of the filter  805  uses NLMS. However, persons of ordinary skill in the art will readily appreciate that the filter  805  could utilize some other suitable filter structure or filtering arrangement; and/or other forms of coefficient adaptation (e.g., LMS, RLS, etc.) could be implemented. It will be readily appreciated by persons of ordinary skill in the art, that equalizer trainer  1275  is also implemented in the example manner of  FIG. 12 . 
     Persons of ordinary skill in the art will readily appreciate that a digital baseband receiver using receive diversity receiving signals from the wireless base station  110  using transmit diversity is readily constructed using appropriate combinations of the example implementations discussed above. For example, if the wireless base station is implementing STTD an example implementation digital baseband receiver would be a combination of  FIGS. 9 and 12 . In particular: four equalizer trainers are needed (2 for transmit and 2 for receive); filter outputs (from two equalizer trainers associated with each transmit antenna) are combined to form a common error signal (as discussed in connection with  FIGS. 12-13 ); an equalized/despread output for each transmit antenna is obtained by summing corresponding outputs from each receive antenna. For the case of STTD, two successive symbols are input to an STTD decoder. For the CLTD case, equalized/despread outputs corresponding to the second transmit antenna (after summing across receive antennas) are multiplied by a complex antenna verified weights and then summed with equalized/despread outputs corresponding to the first transmit antenna. Alternatively, to limit the complexity of the receiver, one may choose to do either transmit or receive diversity but not both simultaneously and, thus, have only two effective equalizers in the receiver. 
     Persons of ordinary skill in the art will also appreciate that a single computing device or processor may be utilized to implement a plurality of functions. For example, when transmit and receive diversity are both present, as discussed above, four equalizers are needed. However, they may not necessarily be implemented as four physical equalizers, but could alternatively, be implemented as two equalizers at twice the clock speed. Further, to limit the complexity of a wireless receiver, the wireless receiver may implement only transmit diversity techniques (e.g.,  FIGS. 9 and 11 ), receive diversity techniques (e.g.,  FIG. 12 ), or neither (e.g.  FIG. 4 ). 
       FIG. 14  is a block diagram illustrating a portion of a disclosed example digital baseband receiver  1400  configurable to implement: (a) improved equalizer training (as discussed above), (b) conventional CPICH based NLMS, or (c) a RAKE receiver. To support all three implementations, the example digital baseband receiver  1400  implements two multiplexers  1405  and  1410 . The first multiplexer  1405  selects whether the equalizer trainer  435  uses (a) the training signal  430  (i.e., multipath pilot signal) or (b) the delayed input chip level signal  452 . The second multiplexer  1410  selects whether the equalizer coefficients  1420  are substantially equivalent to (a) complex conjugates  1425  of channel estimates  412  or (b) the coefficients  455  generated by the equalizer trainer  435 . To form the complex conjugates  1425  of the channel estimates  412 , the receiver  1400  provides a complex conjugator  1435 . The complex conjugator  1435  forms complex conjugates  1425  as a complex conjugate of each channel estimate  412 . 
     It should be readily apparent to persons of ordinary skill in the art that the receiver  1400  could eliminate either of the multiplexers if only two of the three receiver architectures are implemented. 
     A Rake receiver can be realized by directly using the complex conjugates  1425  of the channel estimates  412  as filter coefficients  1420  for the equalizer  445 . In the illustrated example, the equalizer  445  is an FIR filter, and the t K   th  coefficient  1420  of the filter of the equalizer  445  is substantially equal to the channel estimate  412  corresponding to a multipath delay of t K  (see  FIG. 6 ), where K is an index taking on values between 1 and n where n is the number of multipaths processed by channel estimator  405 . Zero values are assigned for equalizer coefficients  1420  corresponding to non-existing multipaths. Because each tap of the filter of the equalizer  445  (i.e., input data flowing through the filter) holds a delayed input data sample which is multiplied by the complex conjugate  1425  of the channel estimate  412  of corresponding multipath, an output  454  of the equalizer  445  is a sum of phase corrected samples at multipath positions. Thus the filter of the equalizer  445  implements processing substantially equivalent to performing maximal ratio combining at a chip level. In the illustrated example, if not all multipath information is available, channel estimates  412  corresponding to available coefficients  1420  of the filter of the equalizer  445  are computed. Then paths with high energy are chosen based on the amplitude of the channel estimates  412 . In the illustrated example: if the amplitude of the channel estimate  412  is larger than a threshold, set the corresponding coefficient  1420  substantial equal to the complex conjugate of the channel estimate  412 ; otherwise set the corresponding coefficient  1420  to zero.  FIGS. 15   a - c  illustrates an example of a 16-tap FIR filter case. In the example of  FIGS. 15   a - c , selected multipaths by a threshold are with delays at 4, 6, 7, 8, 10, and 11. The channel estimates h( 4 ), h( 6 ), h( 7 ), h( 8 ), h( 10 ), and h( 11 ) at these paths provide the filter coefficients h*( 4 ), h*( 6 ), h*( 7 ), h*( 8 ), h*( 10 ), and h*( 11 ) accordingly. Other coefficients are set to zero. 
     The illustrated example of  FIG. 14  can be configured to support a Rake receiver by setting the multiplexer  1410  to select the complex conjugates  1425  (the setting of multiplexer  1405  can be ignored). It can further be configured to support a traditional CPICH based NLMS by setting the multiplexer  1405  to select the delayed input chip level signal  452  and the multiplexer  1410  to select the coefficients  455 . Further, by setting the multiplexer  1405  to select the training signal  430  and the multiplexer  1410  to select the coefficients  455 , the improved equalizer training discussed above is implemented. In the illustrated example, when one of the multiplexers  1405 ,  1410  is set so that the output of the complex conjugator  1435 , the pilot regenerator  415 , the summer  425 , the multiplexer  1405 , and/or the equalizer trainer  435  are ignored it is bypassed or disabled to reduce power consumption. 
     In the illustrated example, the first and second multiplexers  1405  and  1410  are controlled by a processor (not shown). For example, the processor controls the first and second multiplexers  1405  and  1410  to maximize the performance of the wireless receiver. For example, the processor could choose a RAKE receiver for low SNR, high Doppler, and strong line of sight conditions, and choose the improved equalizer training (as discussed above) otherwise. The processor may also control the first and second multiplexers  1405  and  1410  based on user input, wireless base station operator input, or any other suitable criteria. 
     As is well appreciated by persons of ordinary skill in the art, the example of  FIG. 14  can be readily extended (using the teachings of  FIGS. 9-13 ) to support wireless base station transmit diversity and receive diversity for (a) improved equalizer training (as discussed above), (b) conventional CPICH based NLMS, or (c) a RAKE receiver. 
     A flowchart representative of example machine readable instructions that may be executed by the example digital baseband receiver  400  of  FIG. 4  is shown in  FIG. 16 . In this example, the machine readable instructions comprise a program for execution by a processor such as a digital signal processing (DSP) core  1710  shown in an example digital signal processor  1700  discussed below in connection with  FIG. 17 . The program may be embodied in coded instructions stored on a tangible medium such as a CD-ROM, a floppy disk, a hard drive, a digital versatile disk (DVD), or a memory associated with the processor  1710 , but persons of ordinary skill in the art will readily appreciate that the entire program and/or parts thereof could alternatively be executed by a device other than the processor  1710  and/or embodied in firmware or dedicated hardware in a well known manner. For example, any or all of the channel estimator  405 , the pilot regenerator  415 , the summer  425 , the equalizer trainer  435 , the PN generator  410 , the delays  420 , 440 , 450 , the equalizer  445 , and/or the despreader  460  could be implemented by software, hardware, and/or firmware. Further, although the example program is described with reference to the flowchart illustrated in  FIG. 16 , persons of ordinary skill in the art will readily appreciate that many other methods of implementing the example digital baseband receiver  400  of  FIG. 4  may be used. For example, the order of execution of the blocks may be changed, and/or some of the blocks described may be changed, eliminated, or combined. 
     The program of  FIG. 16  begins at block  1605  where the digital baseband processor  225  sets, configures and initializes the analog baseband processor  220  and the RF transceiver  215 . The digital baseband processor  225  detects the signal level associated with received multipaths (block  1610 ). The digital baseband processor  225  next determines the timings associated with the received multipaths (block  1615 ) and sets the receiver mode (e.g., RAKE, improved equalizer training discussed above, etc.) (block  1617 ). Next, the channel estimator  405  despreads the input chip level signal  326  (block  1620 ) and determines channel estimates  412  (block  1625 ). Using the outputs  412  of the channel estimator  405 , the pilot regenerator  1630  and the summer  425  determine the training signal  430  (i.e., multipath pilot signal) (block  1630 ). The equalizer trainer  435  adapts the coefficients  455  based upon the training signal  430  (block  1635 ). Using the adapted coefficients  455 , the equalizer  445  equalizes the delay input chip level signal  452  (block  1640 ). Finally, the despreader  460  despreads the equalized receive signal  454  and the remainder of the receiver processing is applied (block  1645 ). As receiving of the input chip level signal  326  continues, the process comprised of block  1620 - 1645  is repeated. 
       FIG. 17  is a block diagram of an example digital signal processor (DSP)  1700  capable of implementing the apparatus and methods disclosed herein. For example, the DSP  1700  can be implemented by one or more digital signal processors from Texas Instruments. Of course, other digital signal processors from other manufacturers are also appropriate. 
     The DSP  1700  of the instant example includes the DSP core  1710 . The DSP core  1710  is a general purpose programmable processor with enhancements making it more suitable for real-time processing of digital signals. The DSP core  1710  executes coded instructions present in main memory of the DSP  1700 . The DSP core  1710  may implement, among other things, the equalizer trainer  435  and/or the equalizer  445 . 
     The DSP core  1710  is in communication with the main memory including a read only memory (ROM)  1720  and a random access memory (RAM)  1725  via a bus  1705 . The RAM  1725  may be implemented by Synchronous Dynamic Random Access Memory (SDRAM), Dynamic Random Access Memory (DRAM), RAMBUS Dynamic Random Access Memory (RDRAM) and/or any other type of random access memory device. The ROM  1720  may be implemented by flash memory and/or any other desired type of memory device. Access to the main memory  1720 ,  1725  is typically controlled by a memory controller (not shown) in a conventional manner. The RAM  1725  may implement, among other things, the delays  420 ,  440 , and  450 . 
     To reduce the computational burden of the DSP core  1710 , the DSP  1700  provides an accelerator  1715 . The accelerator  1715  contains dedicated circuits and hardware to implement specific data manipulation and/or signal processing. The accelerator  1715  may implement, among other things, the channel estimator  405 , the pilot regenerator  415 , the summer  425 , and/or the despreader  460 . 
     The DSP  1700  also includes a conventional interface circuit  1730 . The interface circuit  1730  may be implemented by any type of well known interface standard, such as an external memory interface (EMIF), serial port, general purpose input/output, etc. 
     One or more input devices  1735  are connected to the interface circuit  1730 . The input device(s)  1735  (e.g., analog to digital converters, data buffers, external memory, etc.) may be used to provide the DSP core  1710  input data and signals to be processed. 
     One or more output devices  1740  are also connected to the interface circuit  1730 . The output devices  1740  (e.g., digital to analog converters, data buffers, external memory, etc.) may be used by the DSP core  1710  to provide processed output data and signal to external devices. 
     From the foregoing, persons of ordinary skill in the art will appreciate that the above disclosed methods and apparatus may be realized within a single device or across two cooperating devices, and could be implemented by software, hardware, and/or firmware to implement the improved wireless receiver disclosed herein. 
     Although certain example methods, apparatus and articles of manufacture have been described herein, the scope of coverage of this patent is not limited thereto. On the contrary, this patent covers all methods, apparatus and articles of manufacture fairly falling within the scope of the appended claims either literally or under the doctrine of equivalents.