Abstract:
The present invention relates amplifiers using metal oxide semiconductor based integrated circuits. The present invention is particularly but not exclusively related to audio application mixed signal chips. The present invention provides an analog circuit for processing analog signals in an integrated circuit comprising a number of metal oxide semiconductor transistor devices, the circuit stage comprising a first said transistor device having a first oxide thickness, and a second said transistor device having a second and different oxide thickness. Preferably a cascode based op amp structure is implemented.

Description:
FIELD OF THE INVENTION 
   The present invention relates to amplifiers using metal oxide semiconductor based integrated circuits. The present invention is particularly but not exclusively related to audio application mixed signal chips. 
   BACKGROUND OF THE INVENTION 
   Due to the increasing need for miniaturisation of portable devices such as MP 3  players, mobile telephones and personal digital assistants, it has become increasingly important to implement both digital processing functions and related analogue, especially audio, processing functions on the same chip or integrated circuit—a so called mixed signal chip. For this reason the implementation of analogue functions on metal oxide semiconductor (MOS) based devices has become increasingly important. 
   A major problem with this technology however is that the digital and analogue circuits require different characteristics from the same semiconductor technology. Digital circuits are fastest and consume least power and chip area when implemented using the shortest channel available in a given manufacturing technology. However this limits the supply voltage that can be applied to this digital circuitry without causing breakdown or premature wear-out of the small devices used. For example digital circuitry on a currently mature process might use a structure with 0.35 um drain-source spacing and 70 nm gate oxide thickness. However most analogue circuits must operate in accordance with legacy standards, for example in order to provide a 2V rms signal for consumer standard audio Line Outputs, or possibly 5V rms for professional applications. Operation at these large signal swings is likely to continue to be necessary for some time in the future, to maintain the ratio of signal power to that of the thermal noise of the op amps and resistors in the signal path: a halving of the signal amplitude would require the noise power to be divided by four, reducing the required circuit impedances by a factor of four, and increasing the power required by the amplifiers, despite the lower supply voltage. Also reducing the signal level would increase the relative importance of extraneous noise and interference coupling into the circuitry. This is likely to require a rail to rail supply as high as 18V. To avoid large electric fields in the device, which cause breakdown or long-term reliability issues such as hot-carrier induced degradation of threshold voltage and transconductance, this requires a larger device structure, with typically a 3 um minimum drain-source spacing and a 350 nm oxide layer thickness. 
   Many widely available semiconductor manufacturing technologies today offer the possibility of selecting a thin or thick gate oxide thickness for selected transistors in each integrated circuit on the basis of additional photographic masking and processing steps. For instance one technology allows 70 nm gate oxides for 3.3V nominal (3.6V maximum) operation for core logic transistors, but 120 nm devices for 5V nominal (5.5V maximum) digital input and output devices. Such technology has been used for mixed signal circuits where the logic operates at 3.3V and all the analogue circuitry operates at 5V. Similarly circuits such as LCD display drivers use 3.3V control logic and 18V nominal (19.8V maximum) (350 nm gate oxide) output stages. Allowing the logic to operate at lower voltage using smaller devices makes these devices smaller in chip area and hence cost and reduces the power consumed by the digital circuitry. 
   High performance audio op amps also require high open loop bandwidth so that the distortion inherent in their open-loop transfer characteristics may be suppressed across the audio band by negative feedback around the amplifier, even when this feedback is relatively light, to provide gain in the signal path. Suppression of distortion is important even at frequencies well above the human hearing limit of approximately 20 kHz. This is because distortion at these higher frequencies, for example up to 100 kHz, have an effect in the audible range (20 Hz–20 kHz). Also audio signals from delta-sigma digital-to-analogue converters have quantisation noise components well above the audio band, which can intermodulate to produce audio band components unless the amplifier retains a linear closed-loop response to these high-frequency components. A wide closed-loop bandwidth is also necessary to avoid relative phase delays across the audio spectrum. 
   One of the major sources of noise in MOS technology is flicker noise. MOS devices such as transistors contain traps, due to impurities and inevitable imperfections in the crystal structure of the silicon, in or near the interface between the silicon and silicon oxide layers. The current in MOS devices typically travels substantially along this interface, and the traps charge and discharge randomly over time. This gives rise to a noise component of charge density at the oxide interface with an approximately 1/f power spectrum, i.e. with higher spectral density at lower frequencies. 
   For circuit analysis, this charge variance ΔQ may be regarded as an equivalent modulation of the gate voltage ΔVG where ΔVG=ΔQ/Cox, Cox being the capacitance from gate to channel across the gate oxide. Cox is inversely proportional to gate oxide thickness, so for the same charge variance, the equivalent gate voltage noise is proportional to gate oxide thickness. In practice, this is found to be the case, i.e. similar processes with different gate oxide thicknesses give gate flicker noise voltages increasing with gate oxide thickness. 
   It is also found that the flicker noise voltage is inversely proportional to the square root of the area of a MOS transistor. So one approach to reducing this noise is to increase the surface area, i.e. to increase the width and length of the transistor. However to improve flicker noise by say 6 dB would require four times the transistor area: significant further improvement in flicker noise rapidly leads to impractically large devices, both in terms of extra parasitic capacitances and in the chip area consumed and hence in cost of manufacture. 
   The contribution of flicker noise of a given transistor to the input-referred noise voltage of an amplifier can also be reduced by altering the gain from the transistor referred to the input, by altering its aspect ratio or altering its bias current. But this makes the design deviate from what would otherwise be considered the optimum in terms of the desired combination of area, power, and performance, and in practice there is again only a small improvement practically achievable without unduly compromising other design objectives. 
   Also chopper-stabilisation techniques could be incorporated to move flicker noise away to higher frequencies, where the noise can either be ignored or filtered out. However this adds to the complexity of the circuit, generally requiring the addition of multiple switches and clock generation and distribution circuitry, and tends to give spurious output signals at the chopping frequency and its harmonics. 
   So in general, for a given circuit topology, circuit specification, and manufacturing technology, there is a practical and economical lower limit to the flicker noise achievable. 
   A known circuit common in high performance audio amplifier applications is the differential folded cascode op amp circuit, a schematic for which is shown in  FIG. 1   a . This circuit offers low distortion, high gain, and wide bandwidth, which are desirable for hi fidelity sound reproduction. The operation of such circuits is well known to those skilled in the art, however the cascode arrangement essentially utilises a gain transistor (MP 1  or MP 2 ) together with a cascode transistor (MNC 1  or MNC 2 ) which effectively reduces the variation in voltage across its associated gain transistor (MP 1  or MP 2 ) in order for this to amplify changes in its input voltage in a linear fashion; thus reducing distortion. This topology also offers high voltage gain to the output lout and wide voltage compliance at this node, either for directly driving an output or to act as the input of a further op amp gain stage. 
     FIG. 1   a  shows a differential folded cascode amplifier structure using two cascode transistors (MNC 1 , MNC 2 ) and constant current bias devices MNM 1 , MNM 2 . Since bias device MNM 2  passes a constant current, all signal current from input device MP 2  passes through cascode device MNC 2  to the output lout. Similarly, signal current from MP 1  passes through cascode device MNC 1  rather than bias device MNM 1 , and is then mirrored by mirror devices MPB 1 , MPB 2  to the output lout. Cascode devices MPC 1  and MPC 2  are inserted in series with the drains of MPB 1 , MPB 2  to improve the output impedance and accuracy of this current mirror. Suitable bias voltages VCP 1 , VCN 1 , VBN 1  are derived by other circuitry using standard techniques. 
   The folded cascode structure of  FIG. 1   b  is a variation of this differential folded cascode amplifier. In this case previous bias devices MNM 1  and MNM 2  are reconnected as mirror devices with MNM 1  being drain-gate connected, and cascode device MNC 1  is also drain-gate connected, and previous mirror devices MPB 1  and MPB 2  now operate as constant bias current sources supplied with a suitable bias voltage VBP 1 . As before, signal current from MP 2  flows through cascode device MNC 2  to the output. However signal current from MP 1  can no longer flow through cascode device MNC 1 , since this is now forced to operate at the constant current supplied by MPB 1 , so this signal current now flows though mirror device MNM 1 , where it is mirrored by MNM 2  and thence flows through MNC 2  to the output. 
   Whilst in these structures the flicker noise contribution of the cascode transistors MNC 1 , MNC 2 , MPC 1  and MPC 2  is small, in practical implementations of the circuits of  FIG. 1   a  or  FIG. 1   b , it is found that the flicker noise contributed by MNM 1  and MNM 2  is one of the dominant components of audio frequency noise, with other flicker noise contributed by input devices MP 1  and MP 2  and by MPB 1  and MPB 2 . This is particularly the case for high-voltage (say 18V) circuits where the amplifier is implemented with appropriately thick gate oxide (say 350 nm) MOS devices. As discussed above, the designer soon reaches a practical lower limit for this flicker noise. Yet there is an increasing requirement for lower and lower noise audio circuitry with better and better signal-to-noise ratio, i.e. lower noise and higher signal swings. 
   SUMMARY OF THE INVENTION 
   In general terms the present invention provides an analogue circuit arrangement using MOS based technology which reduces flicker noise by reducing the oxide thickness of selected transistor devices, having a low operating voltage, compared with those required to operate with a larger operating voltage in the same circuit. The lower voltage transistors are typically employed for biasing, constant current sources, and current mirrors, whereas the larger voltage transistors are exposed to the necessary large signal swings for hi fidelity audio operation. The reduced oxide thickness reduces the flicker noise contribution to the circuit from these low voltage transistors, and hence the overall flicker noise of the circuit. The cascode transistor(s) will still require a thicker oxide layer in order to handle the higher voltage level, whereas the other circuit transistors can be implemented with thinner oxide layers by arranging the circuit such that they are only required to handle relatively low operating voltages. 
   Advantageously, this dual transistor oxide thickness arrangement can be utilised in a folded cascode op amp circuit in which the transistors required to handle the larger voltages are the cascode transistors, which because of the nature of cascode circuits have a much reduced flicker noise contribution compared with transistors in other circuit configurations. Thus a number of the non-cascode circuit transistors can have thinner oxide layers in order to further minimise their contributions to the flicker noise of the op amp circuit. 
   A further advantage of this arrangement is that the overall chip size can be reduced because of the replacement of thicker oxide layer transistors by thinner oxide transistors. For a given width and length and operating current and voltage, the thinner gate oxide transistor will have high transconductance and higher output impedance: conversely for a given requirement for transconductance or output impedance, the width and length can be scaled, giving a smaller chip area occupied by the transistor. Of course this scaling will also reduce the improvement in flicker noise, but this is a tradeoff available to the designer. 
   When used in mixed-signal mixed-voltage integrated circuits, where the digital circuitry uses thin-oxide transistors and the analogue circuitry uses thick-oxide transistors, there is no incremental cost, either in tooling or actual manufacturing cost in using thin-oxide transistors of the same structure as used in the digital circuitry in selected locations in the analogue circuitry. 
   In particular in one aspect the present invention provides an analogue circuit for processing analogue signals in an integrated circuit comprising a number of metal oxide semiconductor transistor devices, the circuit stage comprising a first said transistor device having a first oxide thickness, and a second said transistor device having a second and different oxide thickness. 
   Preferably the first transistor device is arranged in use to have an operating voltage below a predetermined level and the second transistor device is arranged such that in use it is not constrained by this predetermined operating voltage level. This allows the first oxide layer to be thinner than the second oxide layer thickness. For example the predetermined operating voltage level is 3.6V and the first oxide thickness is 70 nm. This compares with an example operating voltage of 19.8V and oxide thickness of 350 nm for the second transistor. 
   Preferably the second transistor device forms part of a cascode transistor device circuit within said analogue circuit. Preferably the cascode transistor device circuit is a differential folded cascode op amp circuit. This reduces the flicker noise contribution of the thicker oxide layer transistors. 
   Preferably the thin oxide transistors are employed in the input, bias and constant current sub-circuits of the op amp circuit. 
   The predetermined operating voltage level may be achieved with the use of a clamp circuit for example. 
   Preferably the analogue circuit is integrated in a mixed signal chip, such as a DAC or ADC chip. The circuit may also be utilised in more complex packages such as system on a chip (SoC), or in a MOS based analogue only integrated circuit. 
   There is also provide a method of processing an analogue signal comprising applying the analogue signal to an analogue circuit for processing analogue signals in an integrated circuit comprising a number of metal oxide semiconductor transistor devices, the circuit stage comprising a first said transistor device having a first oxide thickness, and a second said transistor device having a second and different oxide thickness. 
   There is also provide a method of producing an analogue circuit for processing analogue signals in an integrated circuit; the method comprising providing a number of metal oxide semiconductor transistor devices arranged to implement said circuit stage, at least a first said transistor device having a first oxide thickness, and at least a second said transistor device having a second and different oxide thickness. 
   Preferably the integrated circuit is a mixed signal circuit having additionally digital circuits, which preferably use transistors employing the thinner of the two oxide thicknesses 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Embodiments will now be described with reference to the following drawings, by way of example only and without intending to be limiting; in which: 
       FIG. 1   a  shows a known differential folded cascode op amp circuit; 
       FIG. 1   b  shows another known differential folded cascode op amp circuit; 
       FIG. 2  shows a modified differential folded cascode op amp circuit according to an embodiment; 
       FIG. 3  shows a modified differential folded cascode op amp circuit according to a second embodiment; 
       FIG. 4  shows a modified differential folded cascode op amp circuit according to a third embodiment; 
       FIG. 5  shows a modified differential folded cascode op amp circuit according to a fourth embodiment; and 
       FIG. 6  shows a schematic of an integrated analogue circuit comprising two transistor devices having different oxide thicknesses. 
   

   DETAILED DESCRIPTION 
   Referring initially to  FIG. 1   b , a typical audio op amp design is shown which utilises a differential folded cascode arrangement. MOS based transistor devices MP 1  and MP 2  are input transistors. Signal current from MP 2  passes through folded cascode device MNC 2  to the output. Signal current from MP 1  cannot pass through folded cascode device MNC 1 , since current I(MNC 1 ) is equal to the constant current defined by the constant current source MPB 1 . The MP 1  signal current therefore passes through the current mirror formed by MNM 1 , MNM 2  and then through the cascode device MNC 2  to the output. Thus MNM 1 , MNM 2 , MNC 1 , MNC 2  act as a single ended to differential converter, as well as contributing a cascode function to increase output impedance of this transconductance stage. Transistors MPC 1  and MPC 2  serve as cascode devices to bias devices MPB 1  and MPB 2  to increase their effective output impedance, to maintain the high output impedance at Iout and also to improve power supply rejection. 
   The supply voltage for this circuit block is typically 18V, and so all the transistor devices are “thick” oxide devices, typically 350 nm. Generally NMOS devices contribute more flicker noise than PMOS devices, so the main source of flicker noise in this circuit is MNM 1  and MNM 2 . However the other non-cascode devices (MP 1 , MP 2 , MPB 1 , MPB 2 ) also contribute some noise. To first order the signal current from the drain terminals of cascode devices MNC 1 , MNC 2 , MPC 1 , MPC 2  is equal to that into their respective source terminals, so they can not contribute noise. Second order effects due to their non-zero output conductance and the non-zero output conductance of adjacent devices do allow these devices to generate a small output noise contribution, but this is generally negligible compared to contributions of other devices in the circuit. 
   Referring now to  FIG. 2 , a differential folded cascode op amp according to an embodiment is shown. This circuit comprises the same elements as the circuit of  FIG. 1   b , in particular input transistor devices MP 1  and MP 2  which receive the input signal; input bias transistor MPD 1 ; cascode transistors MNC 1  and MNC 2 ; constant current source transistors MPB 1  and MPB 2 , and associated cascode transistors MPC 1  and MPC 2 , as well as current mirror transistors MNM 1  and MNM 2 . The circuit operates in the same manner as the circuit of  FIG. 1   b , however a number of the transistors (circled) have advantageously a thinner oxide layer than the others. 
   In particular, transistors MNM 1 , MNM 2 , MPB 1  and MPB 2  have thin, for example 70 nm, oxide layer thicknesses, whereas the cascode transistors MNC 1 , MNC 2 , MPC 1  and MPC 2 , as well as the input transistors MP 1 , MP 2  and MPD 1  have thick, for example 350 nm, oxide layers thicknesses. The thicker oxide layers allow these transistors to handle the larger operating voltages imposed upon them by the large signal swing requirement. However the thin oxide layer transistors have lower maximum voltages across their drain-source, gate-drain, and gate-source and so can be implemented with thinner oxide layers, thereby reducing their flicker noise contribution to the circuit. 
   Remaining thick oxide transistors MNC 1 , MNC 2 , MPC 1  and MPC 2  are in cascode configurations so contribute little flicker noise, providing a low noise op amp circuit design. In certain applications, the input transistors MP 1  and MP 2  may also be made thin oxide, to reduce their flicker noise contribution if the input signals can be “guaranteed” to be within the predetermined operating voltage level, for example 3.6V for a 70 nm oxide layer thickness. 
   Whilst a differential folded cascode op amp circuit is the preferred arrangement, other amplifier designs can also benefit from the dual transistor device oxide layer thickness approach. For example non-differential, non-folded, and non-cascode amplifier circuits could be implemented using MOS transistor devices having more than one thickness. 
     FIG. 3  shows an alternative differential folded cascode op amp circuit according to an embodiment. This circuit is similar to that of  FIG. 2 , but includes additional circuitry to ensure the operating drain-source, gate-drain, and gate-source voltages for the thin oxide transistors remain below the predetermined voltage level (eg 3.6V) even in overload conditions. Current mirror device MNM 1  is diode connected and so never sees more current than the sum of bias currents I(MPD 1 )+I(MPB 1 ). Therefore its drain-source voltage can be designed to be less than 3.6V. 
   Complementary current mirror device MNM 2  is not diode connected, however its gate-source voltage is equal to that of MNM 1 , and so is again usually limited to 3.6V; thus permitting thin oxide implementation. However in overload conditions where V(INN) is much larger than V(INP), such that I(MP 2 ) is greater than I(MPB 2 ), the source of MNC 2  potentially can rise in voltage giving excessive drain-source voltage on MNM 2 . To overcome this potential problem a clamp circuit is added comprising transistor device MPX, which clamps the drain voltage of MNM 2  to a safe voltage during overload to avoid excessive drain-source or drain-gate voltage. 
   The gate of MPX can be biased to a suitable voltage in a number of ways, for example the bias circuit comprising R 3 , MNN 3 , MPC 3  and MPB 3  shown. I(MPB 3 ) passing through diode connected MNN 3  and resistor R 3  biases the gate in this example. The voltage is chosen such that MPX turns off in normal operation, but clamps the gate bias voltage of MNM 2  low enough to avoid exceeding a predetermined level, given by the maximum recommended operating voltage of the thin oxide transistor, in overload conditions. Note the clamp transistor MPX is normally inactive, so MPX and its bias circuit contribute no noise. 
   MPB 1  and MPB 2  also contribute some flicker noise, albeit less than the NMOS transistors (MNM 1 , MNM 2 ) and so can be advantageously made thin oxide. 
   In circuits where both input voltages could be guaranteed to stay well below 3.6V, MP 1  and MP 2  could also be made thin oxide. However this is generally hard to guarantee under all overload or transient conditions.  FIG. 4  shows a further alternative op amp circuit according to an embodiment. Here additional input transistor devices MPL 1 , MPL 2 , MPE 1  and MPE 2  are added. Diode-connected devices MPL 1  and MPL 2  are connected back-to-back in parallel between the gates of the input transistors MP 1  and MP 2 , to limit the input differential voltage, assuming one or both inputs are driven from a significant source impedance (e.g. from feedback resistors around the op amp). Assuming the input voltage applied to the gates is now constrained to a known range, cascode devices MPE 1  and MPE 2  can be suitably biased and inserted in series with MP 1  and MP 2  to reduce the maximum drain-source voltage across the input transistors MP 1  and MP 2 . 
   MPE 1  and MPE 2  are cascode devices, so will contribute little noise, and clamp devices MPL 1  and MPL 2  will be off, except during short transients, so will not contribute any noise. 
   Note that in addition to requiring that the drain-source, gate-drain, and gate-source voltages are less than the rated maximum voltage, e.g. 3.6V, of the thin-oxide transistors, the drain-bulk and source-bulk voltages must also be limited to the rated maximum voltage. In the circuit of  FIG. 4 , this condition is satisfied since the bulk of MP 1  and MP 2  is connected to the common source. If the bulk connection was to the positive supply, this condition would be violated for low input voltages, and the device might break down from drain to bulk. 
     FIG. 5  shows an enhanced arrangement of  FIG. 3 , to illustrate the use of this technique in a two-stage amplifier, which additionally includes a conventional class A output stage. The class A gain transistor device MN 4  is biased by MPD 4 , with Miller compensation using resistor RC and capacitor CC, as well as a lever shifter MN 3 . 
   These techniques could be equally applied to the amplifier of  FIG. 1   a  with MNM 1 , MNM 2 , MPB 1  and MPB 2  thin oxide. In this case clamps analogous to MPX would need applying to the sources of MNC 1  and MNC 2 . MP 1  and MP 2  could be made thin-oxide with the addition of MPE 1 , MPE 2 , MPL 2 , MPL 1  as above. Similar adaptations could be made to other similar amplifiers. 
   Though the above has been described to a process with 3.6V and 19.8V transistors available, the concept could obviously be applied to processes with other maximum transistor operating voltages with clamps applied to limit the voltages across thin-oxide transistors to other predetermined voltages. 
   These analogue circuit stages can be used as straightforward audio analogue amplifier stages, or combined with digital circuits for example in a digital to analogue converter (DAC) or an analogue to digital converter (ADC) on a mixed signal integrated circuit. In such an arrangement, the same type of thin oxide transistor devices used in the analogue circuit stage may also be implemented in the digital circuit(s). 
   In principle, the oxide thickness of every noise-contributing transistor could be optimised to minimise flicker noise according to its maximum applied voltage. However each option of oxide thickness requires at least one extra photo mask to be tooled, and one extra photolithographic wafer processing step, and one extra oxide growth wafer processing step, so more than say three different oxide thicknesses becomes expensive in tooling costs and wafer processing costs. Typically just two will be adequate: one for voltage-limited devices, and one for devices which may see the full analogue supply voltage. 
     FIG. 6  shows schematically two MOS transistor devices in the same analogue circuit on an integrated circuit, but with different oxide (SiO 2 ) thicknesses. The oxide layers are not to scale and are merely representative of the above described thick and thin transistor devices. Both devices will be based on a substrate, in the example a p-type substrate. This has two deposits of n-type material to form the source and drain parts of each transistor device. The Silicon Oxide (SiO 2 ) layers at this point are typically thinner than at other parts. The electrical contacts are provided at these thinner regions. Similarly the gate is formed with a contact and a thinning of the oxide layer between the two n type regions. 
   There is however a difference between the two devices in that the oxide layer is thicker on one device (typically 350 nm) at the thinnest part, between gate and channel, compared with the other device (70 nm at the thinnest part). 
   The skilled person will recognise that the above-described apparatus and methods may be embodied as processor control code, for example on a carrier medium such as a disk, CD- or DVD-ROM, programmed memory such as read only memory (Firmware), or on a data carrier such as an optical or electrical signal carrier. For many applications embodiments of the invention could be implemented on a FPGA (Field Programmable Gate Array). Thus the code may comprise for example code for setting up or controlling an FPGA. The code may also comprise code for dynamically configuring re-configurable apparatus such as re-programmable transistor arrays. Similarly the code may comprise code for a hardware description language such as Verilog™ or VHDL (Very high speed integrated circuit Hardware Description Language) which could be used as input to analogue circuit synthesis software. Or analog circuit synthesis software could be written or configured to select appropriate transistors in synthesised amplifiers to be thin-oxide. As the skilled person will appreciate, the code may be distributed between a plurality of coupled components in communication with one another. Where appropriate, the embodiments may also be implemented using code running on a field-(re)programmable analogue array or similar device in order to configure analogue hardware. 
   The skilled person will also appreciate that the various embodiments and specific features described with respect to them could be freely combined with the other embodiments or their specifically described features in general accordance with the above teaching. The skilled person will also recognise that various alterations and modifications can be made to specific examples described without departing from the scope of the appended claims.