Abstract:
A multicarrier spread spectrum (MC-SS) technique is disclosed which includes non-linearly modifying the sub-carriers in the receiver. A method ( 600 ) and receiver ( 200, 300 ) for processing an MC-SS signal, a transceiver for MC-SS communications ( 700 ), and an MC-SS radar ( 800 ) are describe.

Description:
BACKGROUND OF THE INVENTION  
       [0001]     1. Field of the Invention  
         [0002]     The present invention relates generally to multi-carrier spread spectrum communications.  
         [0003]     2. Related Art  
         [0004]     Spread spectrum (SS) systems have proven useful in a variety of applications, including wired and wireless communications, ranging, radar, and synchronization. All of these applications may suffer from interference that is ameliorated by the use of SS techniques. SS operates by greatly expanding the frequency spectrum of the information-containing signal. This expansion is controlled by a spreading code. For example, in direct sequence spread spectrum (DS-SS) the information signal is multiplied by a high rate spreading code. The high rate spreading code creates the wide bandwidth transmit signal. Multi-carrier spread spectrum (MC-SS) is an alternative to the conventional DS-SS and frequency hopping spread spectrum (FH-SS) techniques. MC-SS provides a number of distinct advantages over conventional SS. For example, U.S. Pat. No. 5,521,937 issued to Kondo et al. discloses a MC-SS system having resistance to multipath fading and narrow-band interference. MC-SS systems can also provide improved resistance to partial-band interference and jamming.  
         [0005]     The improved performance of MC-SS over conventional SS is obtained by transmitting each symbol simultaneously across several sub-carrier bands, where the signal on each sub-carrier band is a conventional (although possibly lower bandwidth) spread spectrum signal. At the receiver, the signals from each sub-carrier band are processed and combined. It is difficult to combine the sub-carriers and maintain good performance, since the optimum weighting of the sub-carriers depends on the per sub-channel channel gain, interference/jamming statistics, and noise statistics. For example, U.S. Pat. No. 5,521,937 discloses a maximum ratio combiner (MRC). The MRC combines the sub-carriers by estimating the signal to noise ratio (SNR) on each sub-carrier band, linearly scaling the signal from each sub-carrier band proportionally to the SNR of that sub-carrier band, and then summing all the channels. The MRC combiner must, however, estimate the SNR for each sub-carrier band. Any errors in the estimate of the SNR result in degradation of performance relative to an optimal receiver. Accurate estimation of SNR has proven difficult to achieve in practical systems.  
       SUMMARY OF THE INVENTION  
       [0006]     One embodiment of the invention includes a method for processing a spread spectrum signal. The spread spectrum signal includes a plurality of sub-carrier bands, where substantially similar information is encoded in each of the plurality of sub-carrier bands. The method may include receiving the spread spectrum signal at a receiver and demodulating the spread spectrum signal to recover the plurality of sub-carrier bands. The method may also include modifying the plurality of sub-carrier bands with a predetermined non-linear function to form a plurality of modified sub-carrier bands. By using the method, the need for estimating the SNR in the sub-carrier channels may be avoided.  
         [0007]     Additional features and advantages of the invention will be apparent from the detailed description which follows, taken in conjunction with the accompanying drawings, which together illustrate,.by way of example, features of the invention. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0008]     The following drawings illustrate exemplary embodiments for carrying out the invention. Like reference numerals refer to like parts in different views or embodiments of the present invention in the drawings.  
         [0009]      FIG. 1  is a schematic diagram of an embodiment of an MC-SS transmitter in accordance with the present invention;  
         [0010]      FIG. 2  is a schematic diagram of an embodiment of an MC-SS receiver in accordance with the present invention;  
         [0011]      FIG. 3  is a schematic diagram of an alternate embodiment of an MC-SS receiver in accordance with the present invention;  
         [0012]      FIG. 4  is a graph of the performance of the receiver of  FIG. 2  compared to direct sequence spread spectrum and optimal multi-carrier spread spectrum systems;  
         [0013]      FIG. 5  is a graph of the input-output response of an exemplary non-linearity for the MC-SS receivers of  FIGS. 2 and 3 ;  
         [0014]      FIG. 6  is a flow chart of a method for processing a spread-spectrum multi-carrier signal in accordance with an embodiment of the present invention  
         [0015]      FIG. 7  is an illustration of a pair of MC-SS transceivers in accordance with an embodiment of the present invention; and  
         [0016]      FIG. 8  is an illustration of an MC-SS transceiver configured to operate as a radar in accordance with an embodiment of the present invention. 
     
    
     DETAILED DESCRIPTION  
       [0017]     Reference will now be made to the exemplary embodiments illustrated in the drawings, and specific language will be used herein to describe the same. It will nevertheless be understood that no limitation of the scope of the invention is thereby intended. Alterations and further modifications of the inventive features illustrated herein, and additional applications of the principles of the inventions as illustrated herein, which would occur to one skilled in the relevant art and having possession of this disclosure, are to be considered within the scope of the invention.  
         [0018]      FIG. 1  illustrates a schematic diagram of an embodiment of an MC-SS transmitter  100  in accordance with the present invention. The MC-SS transmitter  100  accepts information symbols  102  s(n), where n represents time. The symbols can optionally be forward error correction encoded user information: For example, forward error correction encoding may be applied to user information using a block, convolutional, or turbo code to generate the information symbols. When the information symbols have been forward error correction encoded, the information symbols may include both user information and redundancy added by the forward error correction coding process. In accordance with one embodiment of the present invention, as illustrated in  FIG. 1 , the information symbols may be applied to each of the sub-carriers. Alternately, different information symbols may be applied to each sub-carrier, where the information symbols have been generated from common user information, e.g. by applying different forward error correction encoding for each sub-channel. In either case, the sub-carriers will carry substantially similar information.  
         [0019]     In accordance with another embodiment of the present invention, the information symbols  102  may be phase modulated, for example using M-ary phase shift keying as is known in the art.  
         [0020]     The information symbols  102  are spread by a spreader  104 , and modulated, by a modulator  106 . The spreader may be implemented by multiplying each information symbol by spreading codes  107  (γ i , i=1 . . . N, where N is the number of sub-carriers) using multipliers  108  to produce spread symbols  110 . According to an embodiment of the present invention, the spreading codes may be different for each sub-carrier. According to another embodiment of the present invention the same spreading code may be used for each sub-carrier, in which case a single multiplier may be used to generate the spread symbols for all sub-carriers.  
         [0021]     The modulator  106  may be implemented by multiplying (mixing) the spread symbols  110  by different sub-carrier frequencies  111  (ω i , i=1 . . . N) using multipliers  112  to produce modulated sub-carriers  114 . The sub-carrier frequencies are shown in  FIG. 1  in a complex form, but real sub-carriers (sine or cosine) may also be used. The modulated sub-carriers  114  are then summed by summer  116  to produce a transmit signal  118  (X(n)). Although it is desirable that the sub-carrier frequencies are selected so that the sub-carriers do not overlap in frequency spectrum, this is not essential.  
         [0022]      FIG. 2  illustrates a schematic diagram of an MC-SS receiver  200  for receiving received signal  201 . The received signal  201  (Y(n)) may be the transmit signal  118  with or without noise, interference, and jamming superimposed. Various ways for communicating the transmit signal to the receiver are possible, including wireless and wired channels. For example, the received signal may be received over a radio link. The received signal may alternately be received through an optical fiber system or over a wire. The channel may cause distortion to the signal, resulting in different sub-channel gain h i  for each of the sub-carrier bands.  
         [0023]     The received signal  201  is separated into sub-carrier bands by a demodulator  202 . According to an embodiment of the present invention, the demodulator  202  may be implemented by multiplying the received signal  201  by the complex conjugate of the sub-carrier frequencies  206  using multipliers  208  to produce demodulated sub-carrier bands  210 .  
         [0024]     The demodulated sub-carrier bands can be processed by non-linearity  311  to produce modified sub-carrier bands  313 . The modified sub-carrier bands  313  can then be despread by despreader  204  and combined by summer  218 . It is not essential that the form of the non-linearity  311  is identical for all N sub-channels. Further discussion of the non-linearity is provided below.  
         [0025]     According to one embodiment of the present invention, the despreader may be implemented by multiplying the modified sub-carrier bands  313  by the conjugate spreading codes  212  (γ i *, i=1 . . . N) using multipliers  214  to produce sub-channel soft symbols.  216 . The despreader may also include the conjugate channel gains h i * in the multiplication. Alternately, multiplication by the conjugate channel gains be performed by a separate multiplier (not shown). Multiplication by the conjugate channel gains may serve to phase align the sub-channels as is discussed further below.  
         [0026]     The soft symbols are referred to as such because they represent a tentative estimate of the information symbols at the transmitter. Rather than making a “hard” (final) decision of the information symbol, a “soft” (tentative) estimate of the information symbols is made, carrying a confidence associated with it (e.g., a weighted value between zero and one is assigned, where zero represents no confidence and one represents highest confidence).  
         [0027]     The conjugate spreading codes may be formed by taking the complex conjugate of the spreading codes  107  (γ i ). Generally, the spreading codes are chosen to have a magnitude of 1 hence, the despreader removes the spreading (since the product of the spreading code by its complex conjugate is equal to one). Of course, various other spreading codes may be chosen, and the spreader and despreader modified accordingly as will occur to one of skill in the art.  
         [0028]     The despreading may be performed at various other points in the receiver. For example, in accordance with an embodiment of the present invention, the multiplication by conjugate spreading codes  212  may be combined with the demodulator  202 .  
         [0029]     In accordance with another embodiment of the present invention, the spreading codes γ i  may be chosen to be the same for all sub-carriers (i.e., γ i =γ for i=1 . . . N). In such a case, a rearranged receiver  300  may be used as illustrated in  FIG. 3 . In the rearranged receiver, the received signal  201  can be fed to despreader  204  to produce a despread signal  304 . The despreader  204  may be implemented with a multiplier  314  to multiply the received signal by the conjugate spreading code  312 . The despread signal can be fed into the demodulator  202  to produce demodulated sub-carrier bands  210 .  
         [0030]     The demodulated sub-carrier bands  210  may be fed through non-linearity  311  to produce modified sub-carrier bands  313 . Sub-channel gains and phase alignment may be compensated for by multiplying the modified sub-carrier bands by the conjugate sub-channel gains h i * using multipliers  318  to produce sub-channel soft symbols  216 . The sub-channel soft symbols  216  can be summed with summer  218  to produce soft symbols  220 . The soft symbols  220  may optionally undergo further processing, such as forward error correction decoding, according to other embodiments consistent with the present invention.  
         [0031]     The non-linearity  311  eliminates the need for a sub-channel signal to noise ratio estimator as required by prior art MRC detectors. Although the non-linearity  311  may be omitted, such a receiver may only perform well if any noise and jamming/interference that may be present is uniform across the sub-channels. The non-linearity  311  can provide a performance improvement relative to a receiver omitting the non-linearity by suppressing jamming and interference signals. For example, when partial-band jamming is present, sub-carrier bands corrupted by jamming may have larger amplitudes than uncorrupted sub-carrier bands due to the additional noise caused by the jamming. The corrupted sub-carrier soft symbols  216  may thus dominate the resulting soft symbols  220 . which can lead to reception errors. To compensate for partial-band jamming the non-linearity  311  can substantially equalize the amplitude of the sub-carrier bands  316 , thus reducing the effects of partial band jamming. Since the modified sub-carrier bands can be approximately equal in magnitude, the jamming corrupted soft-symbols may not dominate the sum, reducing the likelihood of errors in the soft symbols  320 .  
         [0032]     The non-linearity  311  may also provide a similar performance improvement when the noise levels present in sub-carrier bands differ from each other. The non-linearity can reduce the influence of high noise sub-carriers and increase the influence of low noise sub-carriers on the resulting soft symbols  220 , resulting in reduced likelihood of reception errors. Estimation of sub-channel SNR, as required by prior art MC-SS systems is therefore not required.  
         [0033]     Non-linearity  311  may be implemented as an amplitude normalizer, according to an embodiment of the present invention. The amplitude normalizer can modify the sub-carrier bands  313  y(t) such that 
 
 y ( t )= x ( t )/| x ( t )|  (1) 
 
 where x(t) is the demodulated sub-carrier band  210 . This particular embodiment of non-linearity  311  will now be discussed in mathematical detail to provide further understanding of the present inventive concepts. 
 
         [0034]     The received spread spectrum signal may be modeled as: 
 
 r ( n )= s ( n ) H γ( n )+ v ( n )  (2) 
 
 where s(n) is the information symbol, H is a diagonal matrix with the channel gains for different sub-carriers, γ(n) is a spreading vector, and v(n) is a vector of sub-channel noise plus interference/jammer, each at time n. Scalar variables are denoted by lower-case non-bold letters; lower-case bold is used to denote column vectors; and matrices are denoted by upper-case bold. The i th  element of a vector x is denoted by x i.  
 
         [0035]     The spreading vector, y(n) is a vector of the spreading codes, comprised of the spreading codes r, for each sub-channel, i =1 . . . N, γ( n )=[γ 1 ( n )γ 2 ( n ) . . . γ N ( n )] T , where a superscript T denotes the transpose operator. As noted above, the spreading codes may be identical for all the sub-channels, or different spread codes may be used for some or all of the channels.  
         [0036]     Equation (2) may be rearranged to form 
 
 r ( n )= s ( n ) u+v′ ( n )  (3) 
 
 where u is a vector of length N with elements of 1, and 
 
 r ′( n )=( H Γ( n )) −1    r ( n ),  (4) 
 
 v ′( n )=( H Γ( n )) −1   v ′( n ), and   (5) 
 
 Γ(n) is a diagonal matrix whose diagonal elements are the elements of γ(n). 
 
         [0037]     It can be shown by using a constrained minimization and the method of Lagrange multipliers that an optimized set of weights for combining the sub-carrier bands is given by the weight vector  
               w   o     =       1       u   T     ⁢     R       v   ′     ⁢     v   ′         -   1       ⁢   u       ⁢       R       v   ′     ⁢     v   ′           -   1       ⁢   u             (   6   )             
 
 where R v′v′= E└v′( n )v′ T  ( n ) ┘, where E[] is the statistical expectation operator and a superscript H denotes a Hermitian transpose. 
 
         [0038]     As can be seen from equation (6), the optimal weighting requires knowledge of the statistics of the noise and interference/jammer R v′v′ as well as the sub-channel gains H. Estimation of the sub-channel gains (diagonal elements h i  of matrix H) is generally possible, particularly when the channel gain varies slowly in time. Various methods for estimating the sub-channel gains are known in the art. Estimation of the noise and interference/jammer statistics, on the other hand, is difficult to obtain in most situations. Furthermore, any errors in the estimation of the statistics can result in deviation from optimum performance.  
         [0039]     Near optimal performance may be provided by including the non-linearity  311 . For example, when the non-linearity is an amplitude normalizer, as given by equation (1), the elements of the sub-channels after normalization are given by  
                     r   i     ~     ⁡     (   n   )       =         r   i   ′     ⁡     (   n   )                r   i   ′     ⁡     (   n   )                ,           (   7   )             
 
 and the resulting soft symbols, after combining, are given by  
                   s   ~     subo     ⁡     (   n   )       =       ∑     i   =   0       N   -   1       ⁢            h   i          ⁢         r   i     ~     ⁡     (   n   )                   (   8   )             
 
         [0040]     Upon substituting equations (4) and (7) into (8), yields  
               s   ~     subo     ⁡     (   n   )       =       ∑     i   =   0       N   -   1       ⁢            h   i          ⁢           r   i     ⁡     (   n   )           h   i     ⁢       γ   i     ⁡     (   n   )                      r   i     ⁡     (   n   )           h   i     ⁢       γ   i     ⁡     (   n   )                        ,       
 
 which can be simplified in light of the identity  
                 a   b            a   b            =         ab   *            ab   *            ⁢           ⁢   to   ⁢           ⁢   yield             (   9   )                     s   ~     subo     ⁡     (   n   )       =       ∑     i   =   0       N   -   1       ⁢            h   i          ⁢             r   i     ⁡     (   n   )       ⁢     h   i   *     ⁢         γ   i     ⁡     (   n   )       *                  r   i     ⁡     (   n   )       ⁢     h   i   *     ⁢         γ   i     ⁡     (   n   )       *              .                 (   10             
 
         [0041]     Practically speaking, the multiplication by h* serves to compensate for phase error which may be introduced between the sub-channels by the channel, and the multiplication by γ* serves to remove the spreading introduced by the transmitter. In practice, the receiver may estimate the channel gains, and thus provide a multiplication by an estimated h*. Generally, the spreading codes are known to both the transmitter and receiver, although in some applications the receiver may also estimate the spreading code as well. The division by |r i ( n )h i *γi )*| can provide the normalization (non-linearity). Finally, the weighting by |h i | approximates the weighting by signal to noise ratio the MRC combiner (e.g. equation (6)) would provide, but without the complexity of estimating the SNR.  
         [0042]     Equation (10) may be further simplified as  
                   s   ~     subo     ⁡     (   n   )       =       ∑     i   =   0       N   -   1       ⁢       h   i   *     ⁢         γ   i     ⁡     (   n   )       *     ⁢           r   i     ⁡     (   n   )                r   i     ⁡     (   n   )              .                 (   11   )             
 
 assuming that |γ i|= 1, and noting that |h i |=| h i * |. This embodiment of this invention is illustrated in  FIG. 2 . 
 
         [0043]      FIG. 4  illustrates the performance of the amplitude normalizer. The x axis shows the percentage of the sub-channels which are jammed. The y axis shows the signal to noise-plus-interference ratio (SINR) out of the combiner. The results for the receiver  200  using an amplitude normalizer (soft symbols determined according to equation (10)), is shown along with comparison curves for an optimal combiner (combining with weights according to equation (6)), and a direct sequence spread spectrum system (i.e., without multiple sub-channels). All three systems are assumed to occupy the same bandwidth, and the total power of the jammer is 10 dB above the noise level. It can be seen that the amplitude normalizer provides most of the improvement of the optimum combiner relative to conventional direct sequence spread spectrum, losing only 1 to 2 dB when jammer occupy a small percentage of the SS band. Of course, other operating scenarios will result in differing performance, but losses of only a few dB relative to the optimum combiner have been observed in most cases.  
         [0044]     Various other forms for non-linearity  311  may be used in accordance with the present invention. For example, any non-linearity which provides a normalizing effect may reduce the contribution of high amplitude (e.g. jammed or high noise) sub-channels to the soft symbol, and thus provide a performance benefit similar to that obtained by the amplitude normalizer. For example, the non-linearity may be implemented as a limiter as shown in  FIG. 5 . In accordance with another embodiment of the present invention, the characteristics of the limiter, such as the limiting value, may be adjusted based on the channel gins (cnown or estimated).  
         [0045]     The non-linearity may also be implemented as a logarithmic scaling function, e.g. sing a log amplifier. Alternately, the non-linearity may be implemented as an n th  root scaling fimction, e.g. using a square root amplifier or cascade of such amplifiers. Various other embodiments of non-linearity  311  consistent with the present invention will be apparent to one of ordinary skill in the art and in possession of this disclosure.  
         [0046]     During operation of the receiver, the conjugate spreading codes  212  ( FIG. 2 ) are approximately time synchronized with the transmitted spreading codes  107  ( FIG. 1 ) while accounting for propagation delay to allow recovery of the information symbols. Various approaches for accomplishing code synchronization are known in the art. Once the receiver  300  obtains timing synchronization, it is possible to extract ranging information using various techniques known in the art. Additionally, receiver  300  may be used to extract range information from the sub-channel gains, h i , if known, by using the sub-channel gains as an estimate of the channel frequency response, and performing an inverse transform of the channel frequency response to obtain the channel impulse response and resulting delay. This ranging information may then advantageously be applied in determining distance between the transmitter and receiver. This can be beneficial, for example, for position location. Ranging information can also be used to detect distance or length from a single end measurement (where transmitter and receiver are in the same location), for use in collision avoidance or other radar application and for fault location on a wire.  
         [0047]     According to another embodiment of the present invention, a method for processing a spread spectrum signal is illustrated in flowchart form in  FIG. 6 . The method  600  may be applied to a spread spectrum signal having a plurality of sub-carrier bands with each of the plurality of sub-carrier bands having substantially similar information encoded therein. The method may include receiving  602  the spread spectrum signal at a receiver. The method may also include demodulating  604  the spread spectrum signal. Demodulating the spread spectrum signal may be performed to obtain the plurality of sub-carrier bands. The method may also include modifying  606  at least one of the plurality of sub-carrier bands with a predetermined non-linear function to form a plurality of modified sub-carrier bands. The predetermined non-linear function may reduce the high amplitude ofjammed sub-carrier bands, to reduce the effects of jamming, as discussed above.  
         [0048]     In accordance with another embodiment of the present invention, a transceiver  700  is illustrated in  FIG. 7 . The transceiver may include a transmitter  100  and a receiver  200 . The transceiver may be in communication with another transceiver  700 ′ through a channel  702  (e.g. a wireless or wired channel), or the transceiver may be used as a radar as discussed further below.  
         [0049]     According to another embodiment of the present invention, MC-SS radar  800  may be implemented by combining the transmitter  100  and the receiver  200  into a single unit as illustrated in  FIG. 8 . The transmitted signal  118  is reflected back to the receiver  200  by a reflecting surface or object  802 . Antennas  804 ,  806  may be included on both the transmitter  100  and receiver  200 , or the transmitter and receiver may be coupled directly to a wire  808 . Use of MC-SS radar provides advantages over conventional spread-spectrum radar in that greater immunity to partial band jamming and non-white noise may be obtained. Additionally, MC-SS radar may be less affected by variations in the frequency response of the reflecting surface that could cause disruption relative to conventional spread-spectrum radar. This may allow improved performance in radar applications such as distance estimation and material dielectric measurements.  
         [0050]     Recapitulating to some extent, it has been shown how a non-linearity may be used in a spread spectrum multi-carrier receiver to reduce the effects of partial jamming and noise. The non-linearity may ameliorate the effects of jamming and interference. Unlike optimal combiners of the prior art, such as a maximum ratio combiner, no estimate of the signal to noise ratio or an estimate of the statistics of the noise, interference, orjamming is needed. Hence, a multi-carrier spread spectrum receiver using a non-linearity may be less complex than prior art receivers.  
         [0051]     It is to be understood that the above-referenced arrangements are illustrative of the application for the principles of the present invention. Numerous modifications and alternative arrangements can be devised without departing from the spirit and scope of the present invention while the present invention has been shown in the drawings and described above in connection with the exemplary embodiments of the invention. It will be apparent to those of ordinary skill in the art that numerous modifications can be made without departing from the principles and concepts of the invention as set forth in the claims.