Abstract:
A receiver suitable for use in a radio-telephone system for recovering data from encoded, quadrature-modulated communication signals employs a feedback arrangement for suppressing cross-talk between the receiver&#39;s in-phase and quadrature channels to improve data recovery rates during decoding. The receiver converts in-coming analog communication signals into in-phase and quadrature digital signals, which may have cross-talk components. The receiver has an attenuator for subtracting feedback signals from the in-phase and quadrature digital signals to produce cross-talk-attenuated in-phase and quadrature digital signals, a decoder for decoding the cross-talk-attenuated in-phase and quadrature digital signals to generate first and second data output signals, and the above-mentioned feedback arrangement, which preferably employs a recoder and cross-correlation techniques, for generating the feedback signals.

Description:
FIELD OF THE INVENTION 
     The invention relates to telecommunication, and more particularly to techniques suitable for use in digital radio-telephone and other data communication applications for optimizing data recovery from quadrature-modulated communication signals. 
     BACKGROUND OF THE INVENTION 
     Advances in cellular radio-telephony have led to a hybrid analog/digital radio-telephone. A transmitter of such a telephone converts digital signals containing control and message information into analog signals for communication. More specifically, the transmitter forms transmission signals by modulating analog carriers with encoded versions of the digital signals. A common encoding scheme is phase-shift keying in which the digital signals are differentially encoded as changes in phase in accordance with an encoding algorithm. 
     A known modulation technique is quadrature modulation, which entails modulating two orthogonally-related sub-carriers (i.e., two analog signals that are 90 degrees apart in phase) with the encoded data. Typically, in quadrature modulation, the digital signals are converted into two parallel bit streams, and encoded as described above. Then, one of the encoded bit streams modulates a first of the sub-carriers, and the other encoded bit stream modulates a second of the sub-carriers. Subsequently, the modulated sub-carriers are added for transmission. The modulated sub-carriers are called the transmit in-phase (&#34;I&#34;) signal, and the transmit quadrature (&#34;Q&#34;) signals. 
     For recovery of digital data from received encoded, quadrature-modulated signals, a radio-telephone receiver employs a quadrature demodulator. For instance, the quadrature demodulator has a pair of mixers, each of which multiplies the received signal with one of two, different signals generated by a local oscillator and having orthogonally related phases, thereby producing baseband signals. The baseband signals are subsequently converted into digital signals and processed (e.g., filtered) along separate circuit paths, called respectively the &#34;I&#34; and &#34;Q&#34; channels. 
     The resulting signals, i.e., the RECEIVE --  I and RECEIVE --  Q signals, are then decoded to data in a decoder using, essentially, the reverse of the encoding algorithm. Ideally, RECEIVE --  I and RECEIVE --  Q are identical to the corresponding encoded bit streams produced by the encoders in the transmitter, in which case the receiver can recover the data accurately. In other words, the receiver can exhibit a &#34;data recovery rate&#34; of 100%. The data recovery rate is the number of correctly identified or recovered bits in a digital signal of preselected length divided by the total number of bits in that signal. 
     While such a receiver appears generally suitable for its intended purposes, its data recovery accuracy will depend on the extent to which the RECEIVE --  I and RECEIVE --  Q signals as supplied to the decoder are corrupted due to phase and/or amplitude distortion. Distortion in these signals can result in data errors: the more extensive the distortion, the lower the data recovery rate. 
     The distortion causes components of the RECEIVE --  I signal to appear in the RECEIVE --  Q signal, and components of the RECEIVE --  Q signal appearing in the RECEIVE --  I signal. These cross-over components are called &#34;cross-talk.&#34; Unfortunately, decoding of signals corrupted with cross-talk can, and often will, result in data recovery errors, and performance degradation ultimately in the receiver. 
     Cross-talk-producing distortion can originate, for example, either during transmission or within the receiver itself. In cellular radio-telephony, for instance, transmission-originated distortion is a propagation effect arising while the communication signal is traversing the air-waves, e.g., due to multi-path fading. 
     Receiver-originated distortion is caused typically by various combinations of contributing factors, many of which are inherent in electronic devices and signal processing, and often are not readily controllable. For instance, potential sources of such distortion are receiver components having non-linear transfer functions (e.g., amplifiers, mixers, and limiters), signal-synchronization errors, impedance mismatch, filter-center-frequency offsets, electronic-device bandwidth tolerances, and oscillator-frequency drift. 
     SUMMARY OF THE INVENTION 
     Briefly, the invention resides in a feedback arrangement of a receiver, by means of which cross-talk between the receiver&#39;s in-phase and quadrature channels is canceled or, at least, substantially suppressed or attenuated. In so doing, the receiver in accordance with the invention can largely avoid the undesirable affects that cross-talk can have on decoding, and thereby can achieve optimal data recovery rates. 
     The receiver has an analog-to-digital converter for converting an in-coming encoded, quadrature-modulator, analog communication signal into in-phase and quadrature digital signals, each of which may have cross-talk components. The receiver also has an attenuator for subtracting feedback signals from the in-phase and quadrature digital signals to produce cross-talk-attenuated in-phase and quadrature digital signals, a decoder for decoding the cross-talk-attenuated in-phase and quadrature digital signals to generate first and second data output signals, and the above-mentioned feedback arrangement, which preferably employs a recoder for generating a reference signal, and cross-correlation techniques, for generating the feedback signals. 
     The invention is applicable to receivers of radio-telephones as well as of computer and other data communication systems. The invention will find particular utility, however, in cellular radio-telephones. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The aforementioned aspects, features and advantages of the invention, as well as others, are explained in the following description taken in connection with the accompanying drawings, wherein: 
     FIG. 1 is a block diagram of a receiver of an analog/digital communication system in accordance with a preferred embodiment of the invention; and 
     FIG. 2 is a block diagram of a receiver for an analog/digital communication system in accordance with an alternative embodiment of the invention. 
    
    
     DESCRIPTION OF PREFERRED EMBODIMENT 
     FIG. 1 shows a digital receiver 10, which receives communication signals through a communication channel interface 12. For radio telephony, the communication channel interface 12 includes an antenna and often, an amplifier and front-end mixer for producing an intermediate frequency signal. In a computer telecommunication system, the communication channel interface 12 typically includes a port for connection to a communication cable and, often, an amplifier. The separate components making up the communication channel interface 12 are not shown in the drawing. 
     A conventional frequency down-conversion arrangement 14 converts the output of the communication channel interface 12 into two, orthogonally-related signals, sometimes called &#34;in-phase and quadrature baseband signals.&#34; The frequency down-conversion arrangement 14 includes first and second mixers 16, 18, and a local oscillator 20. The local oscillator 20 produces first and second analog signals that differ in phase by 90 degrees. 
     The first mixer 16 mixes the communication channel interface output with the first analog signal from the local oscillator 20 to produce one of the baseband signals, which is then applied to an I-channel 22, and, for that reason, is called I --  SIG. Analogously, the second mixer 18 mixes the communication channel interface output with the second analog signal from the local oscillator 20 to produce the other baseband signal, which is then applied to a Q-channel 24 and is called Q --  SIG. 
     I --  SIG and Q --  SIG are processed in the respective I- and Q- channels 22 and 24 by low-pass filters 26, 28, analog-to-digital (&#34;A/D&#34;) converters 32, 34, and digital filters 36, 38. The low-pass filters 26, 28 typically perform anti-aliasing, and produce band-limited signals for sampling by the A/D converters 32, 34. The digital filters 36, 38 filter the digitized signals from the A/D converters 32, 34 in order to achieve, e.g., adjacent-channel rejection. The output signals from the digital filters 36, 38 are I --  CHAN and Q --  CHAN, respectively. 
     I --  CHAN and Q --  CHAN are applied to respective attenuators 42, 44, which subtract feedback signals therefrom to cancel (or, at least, largely suppress) cross-talk in those signals, and thereby yield substantially uncorrupted RECEIVE --  I and RECEIVE --  Q signals. The generation of the feedback signals will be described in detail below. 
     A conventional decoder 46 is coupled to the attenuators 42, 44 to decode RECEIVE --  I and RECEIVE --  Q, and produce output signals DATA --  OUT1 and DATA --  OUT2. The decoder 46 can generally accurately decode these signals, even in the presence of low-level interference, since the decoder 46 preferably makes decoding decisions through a phase-mapping process which is impervious to such noise. However, when RECEIVE --  I and RECEIVE --  O are corrupted with cross-talk, the phase-mapping process in the decoder 46 can produce bit errors in DATA --  OUT1, DATA --  OUT2. 
     The decoder 46 applies DATA --  OUT1, DATA --  OUT2, to respective output lines 52, 54. DATA --  OUT1, DATA --  OUT2 are subsequently processed, e.g., interleaved in an interleaver 56 to generate a single, output data signal, DATA --  OUT. 
     An output interface 58 connects the receiver 10 to a device (not shown), which is to receive DATA --  OUT. In a radio-telephone system the output interface 58 includes, for example, a voice decoder (i.e., a &#34;vocoder&#34;) and is connected to a speaker. In a computer telecommunication system, the output interface can be connected directly to a computer terminal. The separate components of the output interface 58 are not shown in the drawing. 
     Mathematical Description of Cross-talk 
     Before continuing on with a description of an arrangement for generating the feedback signals, we will now provide a brief mathematical description of the signals processed in the receiver 10 in order to convey a better understanding of the nature of cross-talk, and the advantages of suppressing cross-talk in accordance with the invention. 
     Referring still to FIG. 1, the signal applied to the mixers 16, 18 can be expressed in complex mathematical notation as: 
     
         y(t)=Σ(a.sub.n +jb.sub.n) h(t-τ) EXP{j[2 f.sub.c t+θ.sub.I ]}                                                        EQ (1) 
    
     where &#34;a n  &#34; and &#34;b n  &#34; are the data sequences in the in-phase and quadrature channels, respectively; &#34;t&#34; is time; &#34;τ&#34; is the symbol period; &#34;h&#34; is the pulse response of the channel up to the mixers 16, 18, &#34;f c  &#34; is the carrier frequency; &#34;θ I  &#34; is the carrier phase; and &#34;j&#34; is the square root of (-1), thus denoting an imaginary number. 
     The local oscillator outputs Z(t), jZ(t) are given by Equations (2) and (3), respectively. (The fact that signals Z(t) and jZ(t) are 90 degrees out of phase with one another is represented mathematically by the coefficient, &#34;j&#34;). 
     
         Z(t=EXP [-j(2 ft+θ.sub.II)]                          EQ (2) 
    
     
         jZ(t)=jEXP [-j(2 ft+θ.sub.II)]                       EQ (3) 
    
     where &#34;f&#34; is the local oscillator frequency, and &#34;θ II  &#34; is the local oscillator phase. 
     Mixers 16, 18 mix y(t) with both Z(t) and jZ(t) to yield, respectively, I --  SIG and Q --  SIG, the baseband signals, which are given in Equations (4) and (5). 
     
         I.sub.-- SIG=Σ(a.sub.n +jb.sub.n) h.sub.1 (t-τ)  EQ (4) 
    
     
         Q.sub.-- SIG=jΣ(a.sub.n +jb.sub.n) h.sub.1 (t-τ) EQ (5) 
    
     where h 1  is the pulse response of the channel through to, and including, the mixers 16, 18. 
     As can be seen, I --  SIG includes a factor containing &#34;b n  &#34;, which is the data sequence belonging in the Q-channel; and vice versa, Q --  SIG includes a factor containing &#34;a n  &#34;, which is the data sequence belonging in the I-channel. These cross-over terms, i.e., b n  in I-SIG and a n  in Q-SIG, are cross-talk. 
     This will come further into focus by considering the following: If the pulse response of the channel through to and including the mixers 16, 18 were equal to a real function, e.g., &#34;x,&#34; then Equations (4) and (5) would simplify to Σ[a n  (x)] and Σ[-b n  (x)], respectively. Notice that these terms do not include any cross-talk. 
     On the other hand, if h 1  is a complex number represented by &#34;x+jy&#34;, then, after substituting, Equations 4 and 5 can be rewritten as Equations 6 and 7. 
     
         I.sub.-- SIG=Σ(a.sub.n +jb.sub.n)(x+jy) 
    
     EQ (6) 
     
         Q.sub.-- SIG=jΣ(a.sub.n +jb.sub.n)(x+jy)             EQ (7) 
    
     The real components of these equations is of a special interest. The real component of Equation (6) is given by Equation (8), and the real component of Equation (7) is given by Equation (9). 
     
         Re (I.sub.-- SIG)=Σ(a.sub.n x-b.sub.n y)             EQ (8) 
    
     
         Re (Q.sub.-- SIG)=Σ(b.sub.n x-a.sub.n y)             EQ (9) 
    
     From a practical standpoint, it is the cross-over terms in these real components of I --  SIG and Q --  SIG as given in Equations (8) and (9) that represent detectable cross-talk at the outputs of the mixers 16, 18. 
     Since, as explained above, it would be desirable to cancel the cross-talk from the signals supplied to the decoder 46 in order to optimize the data recovery rate, a feedback arrangement employing a correlation technique for substantially canceling cross-talk will now be described. 
     Feedback Arrangement 
     In accordance with the invention, a feedback-signals-generation sub-circuit 70 generates the feedback signals that the attenuators 42, 44 use to substantially cancel cross-talk in I --  CHAN and Q --  CHAN, respectively. The feedback-signals-generation sub-circuit 70 has a recoder 72, and cross-talk equalizers 74, 76. 
     The recoder 72 is coupled to the decoder 46 to reencode DATA --  OUT1 and DATA --  OUT2 using essentially the identical encoding algorithm to that used in the transmitter (not shown) for encoding the data prior to transmission. The regenerated encoded output signals from the recoder 72 are called I --  RECODE and Q --  RECODE. I --  RECODE and Q --  RECODE are essentially free of any transmission-originated and receiver-originated distortion, unlike RECEIVE --  I and RECEIVE --  Q. These output signals are applied to an I-signal path and a Q-signal path containing the respective I-path cross-talk equalizer 74, and Q-path cross-talk equalizer 76. 
     Each of the cross-talk equalizers 74, 76 has an attenuator 82, 84, a delay 86, 88, first multipliers 92, 94, integrators 96, 98, and second multipliers 102, 104. 
     The I-path attenuator 82 subtracts I --  RECODE from a delayed version of RECEIVE --  I (which is also provided to the decoder 46 as described above). The delay in the supplied RECEIVE --  I is introduced by the delay element 86 and is essentially equal to the inherent delay introduced into I --  RECODE (with respect to RECEIVE --  I) by its signal path including the decoder 46 and the recoder 72. 
     Analogously, the Q-path attenuator 84 subtracts Q --  RECODE from a delayed version of the RECEIVE --  Q signal (which is also provided to the decoder 46 as described above). The delay in the supplied RECEIVE --  Q is introduced by the delay element 88 and is essentially equal to the inherent delay introduced into Q --  RECODE (with respect to RECEIVE --  Q) by its signal path including the decoder 46 and the recoder 72. 
     The first multipliers 92, 94 and integrators 96, 98 together perform cross-correlation functions. The I-path first multiplier 92 multiplies the output of attenuator 82 by Q --  RECODE, and the Q-signal path first multiplier 94 multiplies the output of attenuator 84 by I --  RECODE. The output signals from the first multipliers 92, 94 can be considered &#34;error signals,&#34; which are integrated, i.e., time averaged, in the integrators 96, 98, respectively, to generate cross-correlation factors G IQ , G QI . 
     Where RECEIVE --  I and RECEIVE --  Q have essentially no cross-talk, the input signals to the first multipliers 92, 94 have negligible correlation. On the other hand, where substantial cross-talk is present, the correlation will be higher. The correlation factors G IQ , G QI  represent the extent of correlation in RECEIVE --  I and RECEIVE --  Q. Essentially, therefore, the correlation factors G IQ , G QI  are an indication of the amount of correlated quadrature components in RECEIVE --  I and RECEIVE --  Q. Consequently, the first multipliers 92, 94 and their associated integrators 96, 98 can be regarded as cross-talk correlators. 
     The second multipliers 102, 104 multiply the cross-correlation factors G IQ , G QI  by the respective I-signal and Q-signal outputs of the digital filters 36, 38 to produce the I-path and Q-path feedback signals that are applied to attenuators 42, 44, as described above. 
     Accordingly, the cross-talk equalizers 74, 76 cause the attenuators 42, 44 to substantially cancel cross-talk (or, at least, reduce cross-talk to acceptable limits) by subtracting from I --  CHAN and Q --  CHAN a portion of the orthogonal components of Q --  CHAN and I --  CHAN, respectively. The portions that are subtracted are controlled by the cross-correlation factors G IQ , G QI  via second multipliers 102, 104. Thus, the invention employs a feedback loop to suppress cross-talk between the I and Q channels 22 24 and thereby improve the data recovery rate. 
     FIG. 2 shows a receiver 200 in accordance with an alternative embodiment of the invention. For convenience, the illustrated components of receiver 200 bear the same reference numbers as their counterparts in FIG. 1, and perform the same functions, except for the cross-talk equalizers 202, 204. 
     The essential difference between cross-talk equalizers 74, 76 of FIG. 1 and cross-talk equalizers 202, 204 of FIG. 2, is that the latter does not employ attenuators 82, 84 (FIG. 1). The delayed RECEIVE --  I and RECEIVE --  Q signals from delays 86, 88 are instead applied directly to multipliers 92, 94, where they are multiplied by Q --  RECODE and I --  RECODE, respectively, and then the products are integrated as in the earlier embodiment. 
     Accordingly, cross-talk equalizers 202, 204 of FIG. 2 form the cross-correlation factors by integrating the products of a delayed version of RECEIVE --  I and Q --  RECODE and of a delayed version of RECEIVE --  Q and I --  RECODE, respectively. In contrast, the cross-talk equalizers 74, 76 of FIG. 1 form the cross-correlation factors by integrating the products of Q --  RECODE and the difference between I --  RECODE and a delayed version of RECEIVE --  I, and of I --  RECODE and the difference between Q --  RECODE and a delayed version of RECEIVE --  Q, respectively. It can be expected that the products produced in cross-talk equalizers 74, 76 will typically be smaller than those produced in cross-talk equalizers 202, 204. 
     Therefore, in many applications, cross-talk equalizers 202, 204 will require longer time constants for performing integrations than that required by cross-talk equalizers 202, 204. However, receiver 200 will find potential applications where less cross-talk is typically present, or where timing constraints are less stringent. 
     The foregoing description has been limited to specific embodiments of this invention. It will be apparent, however, that variations and modifications may be made to the invention, while continuing to attain some or all of its advantages. Therefore, it is the object of the appended claims to cover all such variations and modifications as come within the true spirit and scope of the invention.