Abstract:
A complex frequency shift keyed homodyned diplexed radar system and method that can accurately determine the range of one or more targets where the targets have little or no velocity relative to the radar system. The system and method generates a FSK electromagnetic wave that is reflected off the one or more targets and converted into a delayed or phase shifted baseband signal and undelayed baseband signal where the delayed and an undelayed baseband signal may be analyzed to determine the range of the one or more targets.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    This invention relates to radar systems (and sonar and ladar) and methods for determining the range of objects, and more particularly to radar systems and methods for accurately determining the range of objects having little or no relative velocity.  
           [0003]    2. Description of Related Art  
           [0004]    Radio Detection and Ranging (“Radar”) is commonly employed to detect and determine the range of objects or targets relative to the radar system. FIG. 1 is a diagram of a general radar system  1  and a channel or medium  2  that includes a target  30 . As shown in FIG. 1, the radar system includes a transmitter  10  having a transmit antenna  12  and a receiver  20  having a receive antenna  22 . In simple terms, the transmitter  10  generates a signal s(t) that is converted to an electromagnetic wave  14  by the transmit antenna  12 . The signal travels at the speed of light, c away from the transmit antenna  12  in the medium of the channel  2 . The signal may reflect off targets or objects such as the target  30  in the channel  2 . The receive antenna  22  receives the reflected electromagnetic waves and generates a signal s r (t), which is processed by the receiver  20 . It is noted that the transmit antenna  12  and the receive antenna  22  may be in close proximity (monostatic radar systems). Alternatively, the transmitter  10  and the receiver  20  may be separated by a large distance (e.g., in bistatic radar systems).  
           [0005]    In radar systems, if s(t) is a pulsed signal, the received signal s r (t) is nominally equal to αs(t−t r ). In such systems, t r  is the round trip delay or the time required for the electromagnetic wave to travel from the radar transmit antenna to the target and back to the receive antenna and α is an amplitude scaling coefficient. In such systems the range of the target is nominally equal to c×t r /2 where c is the speed of light (approximately equal to 3(10 8 )m/s in a vacuum). If the target is moving away from or toward the radar system (i.e., has a non-zero relative velocity), the relative velocity of the target may be determined by calculating the frequency or Doppler shift of s(t). In particular, it is well known that the velocity of the target, v, is nominally equal to −f d *c/f 0  where f d  is the Doppler frequency and f 0  is the frequency of the transmitted wave  14  of s(t). These principles also apply to sonar and ladar (laser-based) target detection and ranging systems. In ladar the velocity of propagation is also the speed of light (the same as for radar). In sonar the velocity of propagation is the speed of sound (which varies with the nature of the medium in the channel).  
           [0006]    Various radar systems and methods are developed to exploit these well-known attributes to measure the range or velocity of targets in different environments. For example, a prior art system  100  that is used to measure the range and velocity of objects is shown in FIG. 2. As is described below in more detail, the radar system  100  is a homodyned frequency shift keyed (“FSK”) diplex radar system. As shown in FIG. 2, the system  100  includes a signal generator or oscillator  101 , a transmit antenna  102 , a transmit coupler  103 , a receive antenna  106 , a mixer  104 , a switch  108 , a dual anti-alias filter  105 , and a signal processor  107 . The signal generator  101  alternately generates two transmit signals: s 1 (t)=Cos((ω o +ω 1 )t−θ 0 ) and s 2 (t)=Cos((ω o −ω 1 )t−θ 0 ). The signal generator  101  is thus a diplexed signal generator that alternates between the generation of the s 1 (t) and s 2 (t) signals. The transmit signals s 1 (t) and s 2 (t) are transmitted by the transmit antenna  102  via the transmit coupler  103 . The receive antenna  106  receives the reflected signals s r (t) from target objects where the signals are in the form of s(t−τ) (switching between s 1 (t−τ) and s 2 (t−τ)). Accordingly, s r (t) is equal to either: Cos((ω o +ω 1 )(t−τ)−θ 0 ) or Cos((ω o −ω 1 )(t−τ)−θ 0 ).  
           [0007]    The received signal s r (t) and the transmit signals s 1 (t) and s 2 (t) are downconverted (mixed and low-pass-filtered) by the mixer  104  with the “local oscillator” (“LO”) signal Cos((ω o +ω 1 )t) and Cos((ω o −ω 1 )t). The variable θ 0  represents the phase delay of the signal between the transmit antenna  102  and the mixer  104  LO signal. The resultant signal is the low pass filter (“LPF”) of s r (t)x s 1 (t) or s 2 (t), which is either:  
             LPF {Cos((ω o +ω 1 ) t ) Cos((ω o +ω 1 )( t −τ)−θ 0 )}=Cos((ω o +ω 1 )τ+θ 0 )  Eq. 1  
             LPF {Cos((ω o −ω 1 ) t ) Cos((ω o −ω 1 )( t −τ)−θ 0 )}=Cos((ω o −ω 1 )τ+θ 0 ).  Eq. 2  
           [0008]    The switch  108  is synchronized to the changes in frequency at the diplexed transmit signal generator  101  and thus generates two different outputs at ports  110  and  112  having signals, F1 and F2 nominally equal to Eq. 1 and Eq. 2 after anti-alias filtering by the dual anti-alias filter  105 .  
           [0009]    In the above equations, “τ” is the round trip propagation delay to the target. By substituting τ=(2/c)(R+Vt) and by letting ω d =ω 0 (2V/c) (note that the Doppler frequency is ƒ d =2Vƒ 0 /c), θ 0 ′=ω 0 (2R/c)+θ 0 , ω 1 ′=ω 1 (1−(2V/c))≈ω 1 , then ω 0 τ=ω 0 (2V/c)t+ω 0 (2R/c)+τ 0 =ω d t+θ 0 ′ and ω 1 τ+θ 1 =ω 1 (2V/c)t+ω 1 (2R/c)+θ 1 =ω 1 (2V/c)t+θ 1 +2ω 1 R/c=θ 1 +2ω 1 R/c. Therefore the equations that were written in terms of τ can also be written as:  
             F 2=Cos(ω d    t+θ   0 ′+2ω 1   R/c ))  
           [0010]    and  
           F1=Cos(ω d   t+θ   0 ′−2ω 1 R/c)).  
           [0011]    Thus, the F1 and F2 signals of the radar system  100  have the same amplitude and frequency but have a different phase. The phase difference between the F1 and F2 signals is Δφ=2ω 1 τ=2(2ω 1 R/c)=(4π(2ƒ 1 )R/c). Accordingly for this system  100 , the range R is computed by the signal processor  107  as follows: R=(Δφ)c/(4π(Δƒ)) where Δƒ=2ƒ 1  is commonly called the “deviation frequency”. Targets of the prior art system (real FSK diplex Doppler radar) appear as signals of the form Cos(ω d  t+θ 0 ′−2ω 1 R/c))=Cos(ω 0 (2V/c)t+θ 0 ′−2θ 1 R/c)).  
           [0012]    For outbound targets, i.e., targets with increasing range with time, the Doppler shift ƒ d  is negative. For inbound targets, i.e., targets with decreasing range with time, the Doppler shift ƒ d  is positive. The FFT spectrum for real receivers, however, is always symmetrical about its origin. Specifically, the negative frequency portion of the spectrum is equal to the complex conjugate of the positive frequency portion of the spectrum. It is because of this symmetry that target Doppler signals appearing in any Doppler bin may either be inbound targets or outbound targets, thus there exists a velocity direction ambiguity.  
           [0013]    Since the two halves of the spectrum in real receivers contain essentially the same information it is customary in real receivers to only process target information in only one half of the spectrum, e.g., in the positive frequency portion of the spectrum. In the prior art system  101  the direction ambiguity is resolved by observing the polarity of the measured delta phase. Since it is known that target ranges must always be positive it can be inferred whether the target information corresponds to an inbound or outbound target. It must be pointed out that resolving this ambiguity does not resolve inbound and outbound targets in the sense of having independent measurements. It is a weakness of the prior art system that the information for two targets with the same Doppler frequency, e.g., one inbound at +ƒ d  and one outbound at −ƒ d , will have their information appearing in the same FFT Doppler bin, resulting in a single corrupted measurement. The resulting measurement cannot be independent for each target since there is only one measurement. If it were possible for the Doppler information for each target to appear in separate FFT Doppler bins then the two targets would actually be resolved in the sense of having independent measurements for each target.  
           [0014]    Radars may be utilized in many different applications. In some applications, it may desirable to be able to determine the range of a target that has zero relative velocity. Such a system may be desirable when used in conjunction with a cruise control system in a vehicle or a side-facing radar to detect vehicles in adjacent lanes. Given the equations provided above, it is apparent that the prior radar system  100  is unable to determine the range of a target having zero relative velocity since the phase of the DC Doppler return voltage cannot be measured. In some applications for the radar system  100  this limitation may be undesirable or unacceptable.  
           [0015]    In addition to being unable to determine the range of a target having zero relative velocity, the prior art system  100  also has difficulty determining the range of “fading targets”. A target appears as a fading target to a radar system when the radar signal reflected by the target has multiple reflections off the target such as from different points along the surface of a target. The numerous reflections of the signal that are reflected by the target generate constructive and destructive interference. In particular, the reception of multiple signals reflected from a single target can distort the phase of the received signal. In the prior art system  100  shown in FIG. 2, such a distortion of the phase also distorts or limits the accuracy of range determinations.  
           [0016]    Finally, the prior art system  100  of FIG. 2 may not be able to resolve range ambiguities. Target range is calculated by a phase measurement. All phase measurements are ambiguous in multiples of 360°. Therefore, it is possible for the prior art system  100  to detect a target and calculate its range with a large range ambiguity. Consequently, a need exists for a radar system that can accurately determine the range of targets with little ambiguity.  
         SUMMARY OF THE INVENTION  
         [0017]    The present invention includes a complex frequency shift keyed homodyned diplexed radar system and method that can accurately determine the range of one or more targets where the targets have little or no velocity relative to the radar system. The system and method generates a FSK electromagnetic wave that is reflected off the one or more targets and converted into a delayed or phase shifted baseband signal and an undelayed baseband signal where the delayed and undelayed baseband signal may be analyzed to determine the range of one or more targets.  
           [0018]    In one embodiment, the radar system includes an RF signal generator, a first mixer, a delay circuit and a second mixer. The RF signal generator generates a frequency shifted keyed (FSK) RF signal that is converted to an electromagnetic signal and projected towards one or more targets. The first mixer is coupled to the RF signal generator and mixes a received signal (that is reflected off the one or more targets) and the FSK RF signal to generate a real baseband FSK signal. The delay circuit is coupled to the RF signal generator and delays or phase shifts the FSK RF signal. The second mixer is coupled to the delay circuit and mixes the received signal and the delayed FSK RF signal to generate an imaginary baseband FSK signal. The real baseband FSK signal and the imaginary baseband FSK signal can be used to determine the range of one or more targets where the targets have little or no velocity relative to the radar system.  
           [0019]    The radar system may also include a transmit antenna coupled to the RF signal generator where the transmit antenna converts the FSK RF signal to an electromagnetic wave to be directed towards the one or more targets. The radar system may further include a receive antenna coupled to the first mixer and the second mixer. The receive antenna receives electromagnetic waves reflected off the one or more targets and convert the waves to the received signal. The radar system may further include a first switch coupled to the first mixer. The first switch converts the real FSK signal into a first frequency real signal and a second frequency real signal. The system may also include a second switch coupled to the second mixer. The second switch converts the imaginary FSK signal into a first frequency imaginary signal and a second frequency imaginary signal. It is noted that the delay circuit is ideally a phase shifter that phase shifts the FSK RF signal by 90 degrees.  
           [0020]    The present invention also includes a method of determining the range of one or more targets having no relative velocity. The method includes generating a frequency shifted keyed (FSK) RF signal and converting the FSK RF signal into an electromagnetic wave directed toward the one or more targets. The method also receives electromagnetic waves reflected from the one or more targets and converts the electromagnetic waves into a received signal. The method mixes the received signal and the FSK RF signal to generate a real baseband FSK signal. The method delays the FSK RF signal and mixes the received signal and the delayed FSK RF signal to generate an imaginary baseband FSK signal. Finally, the method analyzes the real baseband FSK signal and the imaginary baseband FSK signal to determine the range of the one or more targets.  
           [0021]    The method may further convert the real FSK signal into a first frequency real signal and a second frequency real signal and convert the imaginary FSK signal into a first frequency imaginary signal and a second frequency imaginary signal. This method then analyzes the first frequency real signal, the second frequency real signal, the first frequency imaginary signal, and the second frequency imaginary signal to determine the range of the one or more targets. As in the system above, the method ideally shifts the phase of the FSK RF signal by 90 degrees.  
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0022]    The objects, advantages, and features of this invention will become readily apparent in view of the following description, when read in conjunction with the accompanying Figures, in which:  
         [0023]    [0023]FIG. 1 is a diagram of an application of a basic radar system in a channel.  
         [0024]    [0024]FIG. 2 is a block diagram of a prior art homodyned frequency shift keyed diplexed radar system.  
         [0025]    [0025]FIG. 3 is a block diagram of a complex homodyned frequency shift keyed diplexed radar system in accordance with the present invention. 
     
    
       [0026]    Like reference numbers and designations in the various drawings refer to like elements.  
       DETAILED DESCRIPTION OF THE INVENTION  
       [0027]    Throughout this description, the preferred embodiment and examples shown should be considered as exemplars, rather than limitations on the present invention.  
         [0028]    [0028]FIG. 3 is a block diagram of an exemplary radar system  200  according to the present invention. The system  200  is a complex homodyned FSK diplexed radar system. As shown in FIG. 3, the system  200  includes a signal generator or oscillator  101 , a transmit antenna  102 , a transmit coupler  103 , a receive antenna  106 , a first mixer  104 , a second mixer  204 , a first switch  108 , a second switch  208 , a delay circuit  206 , a quad anti-alias filter  205 , and a signal processor  207 . The signal generator  101  alternately generates two transmit signals: s 1 (t)=Cos((ω o +ω 1 )t−θ 0 ) and s 2 (t)=Cos((ω o −ω 1 )t−θ 0 ). The signal generator  101  is thus a diplexed signal generator that alternates between the generation of the s 1 (t) and s 2 (t) signals. The transmit signals s 1 (t) and s 2 (t) are transmitted by the transmit antenna  102  via the transmit coupler  103 . The receive antenna  106  receives the reflected signals s r (t) from target objects where the signals are in the form of s(t−τ) (switching between s 1 (t−τ) and s 2 (t−τ)). Accordingly, s r (t) is equal to either: Cos((ω o +ω 1 )(t−τ)−θ 0 ) or Cos((ω o −ω 1 )(t−τ)−θ 0 ).  
         [0029]    The received signal s r (t) is downconverted (mixed and low-pass-filtered) by the mixer  104  with the “local oscillator” (“LO”) signal Cos((ω o +ω 1 )t) and Cos((ω o −ω 1 )t) (s 1 (t) and s 2 (t)). The variable θ 0  represents the phase delay of the signal between the transmit antenna  102  and the mixer  104  LO signal. The resultant signal is the low pass filter (“LPF”) of s r (t)×s 1 (t) or s 2 (t), which is either:  
           LPF {Cos((ω o +ω 1 ) t )Cos((ω o +ω 1 )( t −τ)−θ 0 )}=Cos((ω o +ω 1 )τ+θ 0 )= F 2 real   Eq. 3  
           LPF {Cos((ω o −ω 1 ) t )Cos((ω o −ω 1 )( t −τ)−θ 0 )}=Cos((ω o −ω 1 )τ+θ 0 )= F 1 real   Eq. 4  
         [0030]    The switch  108  is synchronized to the changes in frequency at the diplexed transmit signal generator  101  and thus generates two different outputs at ports  110  and  112  having signals, F1 real  and F2 real  as shown in Eq. 4 and Eq. 3 after anti-alias filtering by the dual anti-alias filter  205 .  
         [0031]    In addition, the local oscillator signals (s 1 (t) and s 2 (t)) are phase shifted by 90 degrees by the delay circuit  206 . It is noted that different configurations are possible to phase shift the signals by 90 degrees. For example, the delay circuit  206  may be placed before the mixer  104 . In addition, a 45-degree phase shifter or delay circuit may be placed before the mixer  104  and the mixer  204 . The received signal s r (t) is downconverted (mixed and low-pass-filtered) by the second mixer  204  with the phase shifted local oscillator signals Sin((ω o +ω 1 )t) and Sin((ω o −ω 1 )t). The resultant signal is the low pass filter (“LPF”) of s r (t)×s 1 (t+90°) or s 2 (t+90°), which is either:  
           LPF {Sin((ω o +ω 1 ) t )Cos((ω o +ω 1 )( t −τ)−θ 0 )}=Sin((ω o +ω 1 )τ+θ 1 )= F 2 imag   Eq. 5  
           LPF {Sin((ω o −ω 1 ) t )Cos((ω o −ω 1 )( t −τ)−θ 0 )}=Sin((ω o −ω 1 )τ+θ 0 )= F 1 imag   Eq. 6  
         [0032]    The second switch  208  is also synchronized to the changes in frequency at the diplexed transmit signal generator  101  and thus generates two different outputs at ports  210  and  212  having signals, F1 imag  and F2 imag  nominally equal to Eq. 6 and Eq. 5 after anti-alias filtering by the dual anti-alias filter  205 .  
         [0033]    In the above equations, “τ” is the round trip propagation delay to the target. By substituting τ=(2/c)(R+Vt) and by letting ω d =ω 0 (2V/c) (note that the Doppler frequency is ƒ d =2Vƒ 0 /c), θ 0 ′=ω 0 (2R/c)+θ 0 , ω 1 ′=ω 1 (1−(2V/c))≈ω 1 , then ω 0 τ+θ 0 =ω 0 (2V/c)t+ω 0 (2R/c)+θ 0 =ω d t+θ 0 ′and ω 1 τ+θ 1 =ω 1 (2V/c)t+ω 1 (2R/c)+θ 1 =ω 1 (2V/c)t+θ 1 +2ω 1 R/c=θ 1 +2ω 1 R/c. Therefore the equations that were written in terms of τ can also be written as:  
           F 2 real =Cos(ω d   t+θ   0 ′+2ω 1   R/c ));  
           F 1 real =Cos(ω d   t  +θ 0 ′−2ω 1   R/c));    
           F 2 imag =Sin(ω d   t+θ   0 ′+2ω 1   R/c )); and  
           F 1 imag =Sin(ω d   t+θ   0 ′−2ω 1   R/c )).  
         [0034]    Thus, the F1 and F2 signals of the radar system  200  have the same amplitude and frequency but have a different phase. The phase difference between the F1 and F2 signals is Δφ=2ω 1 τ=2(2ω 1 R/c)=(4π(2ƒ 1 )R/c). Accordingly for this system  200 , the range R is computed by the signal processor  207  as follows: R=(Δφ)c/(4π(Δƒ)) where Δƒ=2ƒ 1  is commonly called the “deviation frequency”. Targets of the system  200  may appear as signals of the form Cos(ω d t+θ 0 ′−2ω 1 R/c))=Cos((ω 0 (2V/c)t+θ 0 ′−2ω 1 R/c)).  
         [0035]    By generating both an undelayed and a delayed signal from mixers  104  and  204 , the radar system  200 , the signal processor  207  can determine whether a target has a positive relative velocity or a negative relative velocity. In particular, due to the phase reference in the signals, the upper sideband can be distinguished from the lower sideband. Further, the present invention can accurately determine the range of targets that have a Doppler shift about zero, i.e., that have little or no velocity relative to the radar system  200 .  
         [0036]    It is an advantage of the radar system  200  that is can resolve the velocity direction ambiguity of the prior art system. The information for two targets of opposite directions and the same magnitude of Doppler frequency, e.g. one inbound at +ƒ d  and one outbound at −ƒ d  will have their information appearing in separate FFT Doppler bins. This is possible because in a complex receiver the time domain signals have the form:  
           Exp ( j (ω d +θ 0 ′−2ω 1   R/c ))= Exp ( j (ω 0 (2 V/c ) t+θ   0 ′−2ω 1   R/c ))  
         [0037]    where j equals the square root of minus one. As is well known to one of ordinary skill in the art, the information of targets with −ƒ d  Doppler frequency appear in the −ƒ d  FFT Doppler bin independently of the information of targets with +ƒ d  Doppler frequency, which appear in the +ƒ d  FFT Doppler bin. All that is needed here is to show how target signals received by the complex radar system  200  can be expressed as a complex rotating phasor time domain signal as given by the above equation. By using Euler&#39;s identity the complex rotating phasor time domain signal can be expressed as Exp(jX)=Cos(X)+jSin(X) where  
           X =(ω 0 +ω 1 )τ+θ 0 =(ω d τ+θ 0 ′−2ω 1 R/c=ω 0 (2 V/c ) t+θ   0 ′−2ω 1   R/c.    
         [0038]    This condition is satisfied when we form the following lower sideband signals for Exp(j((ω 0 −ω 1 )τ+θ 0 )):Cos(X)=F1 real +F2 imag  and Sin(X)=F2 real −F1 imag .  
         [0039]    Alternatively, we could form the following upper sideband signals for Exp(j((ω 0 +ω 1 )τ+θ 0 )): Cos(X)=F1 real −F2 imag  and Sin(X)=F2 real +F1 imag .  
         [0040]    Notice that all four of the necessary signals, F1 real , F1 imag , F2 real , and F2 imag  are formed by the complex FSB diplex Doppler radar system  200 . Thus the complex form of the present invention can both determine the direction of targets and resolve inbound and outbound targets with the same Doppler frequency, unlike the prior art system. Another benefit of such complex signal processing is that there are twice as many FFT Doppler bins (target resolution cells) with independent target information, for a given FFT length, than with a real receiver. This helps resolve targets that would otherwise collapse into the same FFT bin, improving target resolution and target detectability in applications with low target velocities (such as in cruise control or side facing radar) or applications with lots of target fluctuations (such as in detecting people walking).  
         [0041]    One of skill in the art will appreciate that modifications may be made to these embodiments without departing from the spirit and scope of the invention. Accordingly, it is to be understood that the invention is not to be limited by the specific illustrated embodiments, but only by the scope of the appended claims.