Abstract:
A power supply device including a converter having a switch circuit to which an input voltage is supplied and a coil coupled between the switch circuit and an output end from which an output voltage is output; and a control circuit comparing between a feedback voltage and a reference voltage, and on/off controls the switch circuit according to a comparison result; wherein, the control circuit includes a current gradient detection circuit performs detection of a gradient of a coil current flows thorough the coil during an off period of the switch circuit and generates a slope voltage according to a result of the detection; and an adder circuit performs one of generating the feedback voltage by adding the slope voltage to a voltage according to the output voltage and generating the reference voltage by adding the slope voltage to a standard voltage that is set according to the output voltage.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is based upon and claims the benefit of priority from the prior Japanese Patent Application No. 2009-113915 filed on May 8, 2009, the entire contents of which are incorporated herein by reference. 
     FIELD 
     The present disclosure relates to a power supply device, a control circuit and a method for controlling the power supply device. 
     BACKGROUND 
     As a power supply device that generates an output voltage higher than or lower than an input voltage, a comparator type direct-current to direct-current (DC-DC) converter is known according to U.S. published patent application No. 2007/0120547. For example, a step-down DC-DC converter on/off controls a switch circuit to which an input voltage is supplied and a current that flows through a coil coupled to the switch circuit is smoothed by a smoothing capacitor to generate an output voltage. The output voltage generated in the above described manner includes a ripple voltage (ripple component) caused by the coil current and equivalent series resistance (ESR) of the smoothing capacitor. Thus, the DC-DC converter compares the output voltage and a given standard voltage and turns on the switch circuit when the output voltage becomes lower than the standard voltage due to the ripple component, and thereby controls the output voltage. 
     As described above, the DC-DC converter that generates an output voltage by switching the switching circuit demands stabilization of the output voltage, in other words, an output voltage with fewer ripple components. In order to meet the demand, DC-DC converters that use a smoothing capacitor with small ESR have been developed. 
     However, a feed back system and a control cycle of a switch circuit become unstable when a value of ESR of a smoothing capacitor is made small. As a result, there is a drawback in that a voltage VL is irregularly applied to a coil and a current IL that flows through the coil changes irregularly, and thereby an output voltage Vo fluctuates as illustrated in  FIG. 8 . 
     SUMMARY 
     According to aspects of embodiments, a power supply device includes a converter unit that includes a switch circuit to which an input voltage is supplied and a coil that is coupled between the switch circuit and an output end from which an output voltage is output; and a control circuit that performs a comparison between a feedback voltage and a reference voltage, and on/off controls the switch circuit according to a result of the comparison; wherein, the control circuit includes a current gradient detection circuit that performs detection of a gradient of a coil current that flows thorough the coil during an off period of the switch circuit and generates a slope voltage according to a result of the detection; and an adder circuit that performs one of generating the feedback voltage by adding the slope voltage to a voltage according to the output voltage and generating the reference voltage by adding the slope voltage to a standard voltage that is set according to the output voltage. 
     The object and advantages of the invention will be realized and attained by means of the elements and combinations particularly pointed out in the claims. 
     It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are not restrictive of the invention, as claimed. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a block circuit diagram of a DC-DC converter according to one embodiment in accordance with aspects of the present invention; 
         FIG. 2  is a circuit diagram of a timer circuit in accordance with aspects of the present invention; 
         FIG. 3  is a circuit diagram of a current gradient detection circuit in accordance with aspects of the present invention; 
         FIG. 4  is a timing chart illustrating an operation of the DC-DC converter in  FIG. 1 ; 
         FIG. 5  is a waveform illustrating an operation of the DC-DC converter in  FIG. 1 ; 
         FIG. 6  is a block circuit diagram of an alternative DC-DC converter in accordance with aspects of the present invention; 
         FIG. 7  is a timing chart illustrating an operation of the DC-DC converter in  FIG. 6 ; and 
         FIG. 8  is a waveform illustrating an operation of a conventional DC-DC converter. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     Exemplary embodiments will be explained with reference to accompanying drawings. 
     Hereunder, one embodiment in accordance with aspects of the present invention will be described by referring to  FIGS. 1 to 5 . 
     As illustrated in  FIG. 1 , a DC-DC converter may include a converter unit  11  that generates an output voltage Vo based on an input voltage Vi, and a control circuit  12  that controls the converter unit  11  based on the output voltage Vo. 
     The converter unit  11  may include a transistor T 1 , a transistor T 2 , a coil L 1 , and a capacitor C 1 . The transistor T 1  on a main side and a coil L 1  are serially coupled between an input terminal Pi to which an input voltage Vi is supplied and an output terminal Po from which an output voltage Vo is output. Moreover, the transistor T 1  on the main side and the transistor T 2  on a synchronization side are serially coupled between the input terminal Pi and a power line for supplying a voltage lower than the input voltage Vi. 
     The transistor T 1  on the main side and the transistor T 2  on the synchronization side may be N-channel MOS transistors. A first terminal (drain) of the transistor T 1  is coupled to the input terminal Pi to which the input voltage Vi is supplied and a second terminal (source) of the transistor T 2  is coupled to a second terminal (drain) of the transistor T 2 , and a first terminal (source) of the transistor T 2  is coupled to the power line (a ground according to the embodiment) a potential of which is lower than the input voltage Vi. The control circuit  12  supplies a control signal DH to a control terminal (gate) of the transistor T 1 , and supplies a control signal DL to a control terminal (gate) of the transistor T 2 . 
     The transistor T 1  turns on and off in response to a control signal DH. The transistor T 2  turns on and off in response to a control signal DL. The control circuit  12  generates the control signals DH and DL so that the transistor T 1  on the main side and the transistor T 2  on the synchronization side complementarily turn on and off. In other words, the transistors T 1  and T 2  are one example of a switch circuit. The control circuit  12  may include a switch control circuit that turns on and off the transistor T 1  that provides functions as a switch circuit. 
     A coupling point between the transistors T 1  and T 2  is coupled to a first terminal (an input side terminal) of the coil L 1 , and a second terminal of the coil L 1  is coupled to the output terminal Po. Moreover, a second terminal (an output side terminal) of the coil L 1  is coupled to a first terminal of the smoothing capacitor C 1  and a second terminal of the capacitor C 1  is coupled to a ground. The capacitor C 1  may be included in a smoothing circuit that smoothes an output voltage Vo. 
     When the transistor T 1  on the main side turns on and the transistor T 2  on the synchronization side turns off, a coil current IL according to a difference between an input voltage Vi and an output voltage Vo flows through the coil L 1  and the energy (electric power) is accumulated in the coil L 1 . When the transistor T 1  on the main side turns off and the transistor T 2  on the synchronization side turns on, the coil L 1  discharges the accumulated energy and an inducted current (coil current IL) flows through the coil L 1 . The control circuit  12  adjusts a pulse width of control signals DH and DL based on the fed back output voltage Vo. 
     Hereunder, a configuration of the control circuit according to aspects of the embodiment will be described. 
     The control circuit  12  may include a comparator  21 , an RS-flip flop circuit (RS-FF circuit)  22 , a timer circuit  23 , a driving circuit  24 , driver circuit  25 , a driver circuit  26 , a current gradient detection circuit  27 , an adder circuit  28 , a resistor R 1 , a resistor R 2 , and a standard power supply E 1 . 
     A voltage based on the output voltage Vo is supplied to an inverting input terminal of the comparator  21 . According to aspects of the embodiment, a voltage generated by the resistors R 1  and R 2  is supplied. The output voltage Vo is fed back to a first terminal of the resistor R 1  and a second terminal of the resistor R 1  is coupled to a first terminal of the resistor R 2 , and a second terminal of the resistor R 2  is coupled to a ground. A coupling point between resistor R 1  and resistor R 2  is coupled to an inverting input terminal of the comparator  21 . The resistor R 1  and the resistor R 2  generate a voltage (divided voltage, feedback voltage) VFB that is obtained by dividing the output voltage Vo according to respective resistance values. The value of the feedback voltage VFB corresponds to a ratio of the resistance value of resistor R 1  to that of resistor R 2 , and a potential between the output voltage Vo and the ground. Hence, the resistors R 1  and R 2  may generate a feedback voltage VFB that is proportional to the output voltage Vo. 
     A reference voltage VR 1  is supplied to a non-inverting input terminal of the comparator  21 . The comparator  21  compares the feedback voltage VFB with the reference voltage VR 1 , and generates a signal Vc 1  according to the comparison result. In the embodiment, the comparator  21  generates an H level signal Vc 1  when the feedback voltage VFB is lower than the reference voltage VR 1 , and generates an L level signal Vc 1  when the feedback voltage VFB is higher than the reference voltage VR 1 . The signal Vc 1  is supplied to the RS-FF circuit  22 . 
     A signal Vc 1  is supplied to a set terminal of the RS-FF circuit  22  and a signal S 2  is supplied to a reset terminal of the RS-FF circuit  22 . The RS-FF circuit  22  outputs an H level signal S 1  in response to an H level signal Vc 1 , and outputs an L level signal S 1  in response to an H level signal S 2 . In other words, for the RS-FF circuit  22 , the signal Vc 1  may be a set signal, while the signal S 2  may be a reset signal. The signals S 1  output from the RS-FF circuit  22  is supplied to the timer circuit  23  and the driving circuit  24 . 
     The timer circuit  23  outputs an H level pulse signal S 2  in response to an H level signal S 1  after a given time elapses from a rising timing of the signal S 1 . The given time may be a time that depends on an input voltage Vi and an output voltage Vo. In other words, the timer circuit  23  outputs an H level pulse signal S 2  after a given time elapses depending on the input voltage Vi and the output voltage Vo from the rising timing of the signal S 1 . The above described RS-FF circuit  22  outputs an L level signal S 1  in response to an H level signal S 2 . As a result, the signal S 1  output from the RS-FF circuit  22  becomes an H level during a period that depends on the input voltage Vi and the output voltage Vo. In other words, the timer circuit  23  may determine a pulse width of the signal S 1  output from the RS-FF circuit  22 . 
     An example of a timer circuit  23  will be described by referring to  FIG. 2 . 
     As illustrated in  FIG. 2 , the timer circuit  23  may include an operational amplifier  31  and an operational amplifier  32 , an inverter circuit  33 , a capacitor C 11 , a resistor R 11 , and transistors T 11  to T 14 . 
     An input voltage Vi is supplied to a non-inverting input terminal of the operational amplifier  31 . An inverting input terminal of the operational amplifier  31  is coupled to a first terminal of the resistor R 11 , and a second terminal of the resistor R 11  is coupled to a ground. Moreover, a first terminal of the resistor R 11  is coupled to the transistor T 11 . The transistor T 11  may be an N-channel MOS transistor, having a source coupled to the resistor R 11 , a gate coupled to an output terminal of the operational amplifier  31 , and a drain coupled to the transistor T 12 . 
     A potential difference according to a current that flows through the resistor R 11  and a resistance value is caused between both terminals of the resistor R 11 . The operational amplifier  31  generates a gate voltage of the transistor T 11  so that a potential of a node between the resistor R 11  and the transistor T 11  becomes substantially the same as the input voltage Vi. Thus, a current according to the input voltage Vi may flow through the transistor T 11 . 
     The transistor T 12  may be a P-channel MOS transistor, having a source supplied with a bias voltage VB, a drain coupled to the transistor T 11 , and a gate coupled to a drain of the transistor T 12  and a gate of the transistor T 13 . The transistor T 13  may be a type of a MOS transistor which is substantially the same as that of the transistor T 12 , and a source of which is supplied with a bias voltage VB. Accordingly, the transistor T 12  and the transistor T 13  may be included in a current mirror circuit. The current mirror circuit may flow a current to the transistor T 13  that is proportional to a current that flows through the transistor T 11  according to electrical characteristics of the both transistors T 12  and T 13 . 
     A drain of the transistor T 13  is coupled to a first terminal of the capacitor C 11  and the transistor T 14 , and a second terminal of the capacitor C 11  is coupled to a ground. The transistor T 14  is an N-channel MOS transistor having a source coupled to a ground and a drain coupled to the transistor T 13 , in other words, the first terminal of the capacitor C 11 . In other words, the transistor T 14  may be coupled in parallel with the capacitor C 11 . 
     A signal S 1   x  that is obtained by logically inverting the signal S 1  by the inverter circuit  33  is supplied to a gate of the transistor T 14 . The signal S 1  may be a signal output from the RS-FF circuit  22  illustrated in  FIG. 1 , and the transistor T 11  on the main side is turned on when a signal S 1  is an H level, while the transistor T 11  on the main side 1  is turned off when a signal S 1  is an L level. 
     On the other hand, the transistor T 14  turns on when the signal S 1   x  is an H level, in other words, a signal S 1  is an L level, and turns off when the signal S 1   x  is an L level (the signal S 1  is an H level). The transistor T 13  supplies a current depending on an input voltage Vi to the capacitor C 11 . The turned on transistor T 14  couples both terminals of the capacitor C 11  each other, and thereby the first terminal of the capacitor  11  may become a ground level. When the transistor T 14  is turned off, the capacitor C 11  is charged by a current supplied from the transistor T 13 . As a result, a level of the first terminal of the capacitor C 11  may rise from the ground level depending on the input voltage Vi. 
     In other words, the timer circuit  23  short-circuits the both terminals of the capacitor C 11  when the transistor T 1  on the main side illustrated in  FIG. 1  is turned off, and resets a voltage Vn 1  of a node N 12  to a ground level. Charging of the capacitor C 11  starts when the transistor T 1  is turned on. Consequently, the voltage Vn 1  of the node N 12  may rise depending on an input voltage Vi. 
     The node N  12  is coupled to a non-inverting input terminal of the operational amplifier  32 , and an output voltage Vo is supplied to an inverting terminal of the operational amplifier  32 . The operational amplifier  32  compares a voltage Vn 1  of the node N 12  with an output voltage Vo and outputs a signal S 2  according to the comparison result. As described above, the voltage Vn 1  of the node N 12  changes according to the input voltage Vi. The operational amplifier  32  outputs an L level signal S 2  when the voltage Vn 1  is lower than the output voltage Vo, and outputs an H level signal S 2  when the voltage Vn 1  is higher than the output voltage Vo. The voltage Vn 1  rises when the transistor T 1  on the main side is turned on. Accordingly, a period from when the transistor T 1  is turned on to when an H level signal S 2  is output may depend on the input voltage Vi and the output voltage Vo. 
     A current for charging the capacitor C 11 , in other words, a current that flows through the transistor T 11 , is proportional to the input voltage Vi. As a result, a period from when the transistor T 1  is turned on to when the H level signal S 2  is output is inversely proportional to the input voltage Vi. A period from when the transistor T 1  is turned on to when the H level signal S 2  is output is proportional to the output voltage Vo, because the operational amplifier  32  compares the voltage Vn 1  with the output voltage Vo. In other words, a period in which the transistor turns on (on-period) may inversely proportional to the input voltage Vi, and proportional to the output voltage Vo. A period in which the transistor turns off (off-period) may be inversely proportional to the output voltage Vo. Hence, the control circuit  12  may control a switching frequency so as to be substantially constant. 
     As illustrated in  FIG. 1 , the driving circuit  24  generates control signals SH and SL so that transistors T 1  and T 2  complementarily turn on and off based on the signal S 1 . Dead times may be set in the control signals SH and SL so as not to turn on both transistors T 1  and T 2  substantially simultaneously. In the driver circuit  25 , a bias voltage VB is supplied to a power supply terminal of a high potential side and a power supply terminal of a low potential side is coupled to a node between the transistors T 1  and T 2  of the converter unit  11 . The driver circuit  25  outputs a control signal DH in response to a control signal SH. In the driver circuit  26 , a bias voltage VB is supplied to a power supply terminal of a high potential side and a power supply terminal of a low potential side is coupled to a ground. The driver circuit  26  outputs a control signal DL in response to a control signal SL. 
     According to aspects of the embodiment, the driver circuit  24  may output an H level signal SH and an L level signal SL in response to an H level signal S 1 , and output an L level signal SH and an H level signal SL in response to an L level signal S 1 . The transistor T 1  on the main side may turn on in response to a control signal DH based on an H level signal SH, and turn off in response to a control signal DH based on an L level signal SH. Similarly, the transistor T 2  on the synchronization side may turn on in response to a control signal DL based on an H level signal SL, and turn off in response to a control signal DL based on an L level signal SL. 
     The above described comparator  21  outputs an H level signal Vc 1  when the output voltage Vo becomes lower than the reference voltage VR 1 , and the RS-FF circuit  22  outputs an H level signal S 1  in response to the signal Vc 1 . The timer circuit  23  outputs an H level signal S 2  after a given time elapses from the output of the H level signal S 1 , and the RS-FF circuit  22  outputs an L level signal S 1  in response to the signal S 2 . The driver circuit  24  outputs control signals SH and SL for turning on and off the transistors T 1  and T 2  in response to the signal S 1 . 
     Thus, the control circuit  12  may turn on the transistor T 1  on the main side when the output voltage Vo becomes lower than the reference voltage VR 1 , and turn off the transistor T 2  on the synchronization side. After a given time elapses from turning on the transistor T 1  on the main side, the transistor T 1  on the main side is turned off and the transistor T 2  on the synchronization side may be turned on. When the output voltage Vo becomes lower than the reference voltage VR 1  again, the control circuit  12  may turn on the transistor T 1  on the main side and turn off the transistor T 12  on the synchronization side. 
     In other words, the control circuit  12  may turn on the transistor T 1  on the main side for a given period when the output voltage Vo becomes lower than the reference voltage VR 1  and turn off the transistor T 1  after a given time elapses. A period in which the transistor T 1  on the main side is turned on is assumed to be “on-period”, and a period in which the transistor T 1  is turned off is assumed to be “off-period.” The transistor T 2  is controlled so as to turn on and off complementarily with the transistor T 1 , and may turn off during the “on-period”, while turn on during the “off-period.” 
     The current gradient detection circuit  27  detects a negative gradient of a coil current IL that flows through the coil L 1  and generates a correction voltage VS according to the detected gradient. The adder circuit  28  adds the correction voltage Vs generated by the current gradient detection circuit  27  to the standard voltage VRO of the standard power supply E 1  to generate the above described reference voltage VR 1 . 
     The coil current IL that flows through the coil L 1  increases when the transistor T 1  on the main side is turned on and decreases when the transistor T 2  on the main side is turned off. In other words, the waveform of the coil current IL may change with a positive gradient during an on-period and with a negative gradient during an off-period. The change amount of the coil current IL may be a ripple component in the coil current IL, and the ripple component may change in response to turn on and off of the transistor T 1  on the main side. 
     Therefore, the current gradient detection circuit  27  may detect the ripple component of the coil current IL and generate a correction voltage VS according to the ripple component. The coil current IL may decrease gradually when the transistor T 1  on the main side is turned off as described above. 
     The ripple component of the output voltage Vo may be determined mainly by a coil current that flows through the coil L 1  and a resistance element (equivalent series resistance (ESR)) due to couple the capacitor C 1 . In order to reduce a change amount of the output voltage Vo, in other words, to make a ripple component (amplitude) of the output voltage Vo small, a leak current in the capacitor C 1  may be made small, in other words, the value of the ESR may be may be made small. For example, a laminated ceramic capacitor has a small ESR compared with that of a conventionally used electro-conductive polymer capacitor. Thus, using a capacitor with a small ESR value such as a laminated ceramic capacitor may reduce the ripple component and thereby stabilize the output voltage Vo. 
     The comparator  21  of the control circuit  12  compares a feedback voltage VFB that is obtained by dividing the output voltage Vo by the resistors R 1  and R 2  with the reference voltage VR 1 . A change amount of the feedback voltage VFB supplied to the comparator  21  is smaller than that of the output voltage Vo and the feed back voltage VFB is a substantially constant voltage. Hence, it is desirable that the reference voltage VR 1  be changed. The change amount of the reference voltage VR 1  preferably corresponds to the ripple component of the conventional output voltage Vo. According to aspects of the embodiment, the ripple component of the coil current IL is detected and the reference voltage VR 1  is increased according to the ripple component. In other words, the current gradient detection circuit  27  detects the ripple component in the coil current IL and may generate a correction voltage VS that gradually increases according to the ripple component. 
     The adder circuit  28  generates the above described reference voltage VR 1  by adding the correction voltage VS generated by the current gradient detection circuit  27  to the standard voltage VR 0  of the standard power supply E 1 . The reference voltage VR 1  increases according to the ripple component in the coil current IL during an off-period in which the transistor T 1  on the main side is turned off. 
     The comparator  21  outputs an H level signal Vc 1  when the reference voltage VR 1  becomes higher than the feedback voltage VFB. The RS-FF circuit  22  outputs an H level signal S 1  in response to the signal Vc 1  and as a result the transistor T 1  on the main side is turned on. When the transistor T 1  is turned on, a coil current IL of the coil L 1  increases according to a differential voltage between the input voltage V 1  and the output voltage Vo. An output voltage Vo is generated that is smoothed by charging and discharging the capacitor C 1  according to the coil current IL. 
     For example, when an output voltage Vo is decreased due to, for example, an abrupt load change, a timing in which the output voltage Vo becomes lower than the reference voltage VR 1  advances, and a timing in which an H level signal Vc 1  is output advances. In other words, the off-period of the transistor T 1  is shortened. On the other hand, when the output voltage Vo rises, a timing in which the output voltage Vo becomes lower than the reference voltage VR 1  delays. In other words, the off-period of the transistor T 1  may be extended. 
     By the above described operation, a timing in which the transistor T 1  is turned on is determined based on the result of comparison between the output voltage Vo and the reference voltage VR 1 . Therefore, an on-timing (off-time) may be adjusted based on high and low of the output voltage Vo, and a control is applied so that the output voltage Vo may be close to a given voltage (target voltage) based on the standard voltage VR 0 . In other words, the standard voltage VR 0  may be set according to a target voltage that controls the output voltage Vo. For example, under a stable output voltage Vo, a change amount of a coil current IL during an off-period is a substantially constant value. Thus, as illustrated in  FIG. 5 , the transistors T 1  and T 2  turn on and off with a stable interval (cycle). Accordingly, the ripple component of the coil current IL may be reduced compared with a conventional DC-DC converter, and thereby the output voltage Vo may be stabilized. 
     Hereunder, aspects of a configuration example of the current gradient detection circuit  27  will be described. 
     As illustrated in  FIG. 3 , the current gradient detection circuit  27  may include an operational amplifier  41 , a delay circuit  42 , a switch SW 1 , and a capacitor C 21 . 
     A non-inverting input terminal of the operational amplifier  41  is coupled to an input side terminal of the coil L 1  included in the converter unit  11  illustrated in  FIG. 1 , in other words, a node between the transistors T 11  and T 12 , and a voltage VL at the node (hereunder, called as a coil voltage) is supplied. A first terminal of the switch SW 1  is coupled to the node. A second terminal of the switch SW 1  is coupled to an inverting terminal of the operational amplifier  41  and a first terminal of the capacitor C 21 , and a second terminal of the capacitor C 21  is coupled to a ground. A control terminal of the switch SW 1  is coupled to the delay circuit  42 . 
     A control signal DH for controlling on and off of the transistor T 11  on the main side illustrated in  FIG. 1  is supplied to the delay circuit  42 . As illustrated in  FIG. 4 , the delay circuit  42  outputs an H level signal CS in response to an H level control signal DH, and outputs an L level signal CS in response to an L level control signal DH after delaying a given time from the L level signal DH. The switch SW 1  turns on in response to the H level signal CS and turns off in response to the L level signal. 
     The coil voltage VL is supplied to both terminals of the operational amplifier  41  when the switch SW 1  is turned on. The coil voltage VL is supplied to a first terminal of the capacitor C 21 . Hence, a voltage of the first terminal of the capacitor C 21  may be substantially the same as a voltage of the terminal of the operational amplifier  41 . 
     The coil voltage VL is not supplied to the inverting input terminal of the operational amplifier  41  and the first terminal of the capacitor C 21  when the switch SW 1  is turned off. Consequently, a voltage of the inverting input terminal of the operational amplifier  41  becomes the terminal voltage of the capacitor C 21 , in other words, a voltage immediately before the switch SW 1  is turned off is held by the capacitor C 21 . The voltage held by the capacitor C 21  is assumed to be a hold voltage VLs. 
     The above described delay circuit  42  is provided in order to hold a voltage of one of the two input terminals (according to aspects of the embodiment, the inverting input terminal) of the operational amplifier  41  in the capacitor C 21  when voltages supplied to the two terminals of the operational amplifier  41  are substantially the same value. As described above, the non-inverting input terminal of the operational amplifier  41  is directly coupled to the input side terminal of the coil L 1  illustrated in  FIG. 1 , and the inverting input terminal of the operational amplifier  41  is coupled to input side terminal of the coil L 1  through the switch SW 1 . The capacitor C 21  is coupled to the inverting input terminal of the operational amplifier  41 . The coil voltage VL changes according to switching between the transistors T 1  and T 2 . Therefore, there is a period in which voltage levels at both terminals of the operational amplifier  41  are different from each other when the switch SW 1  is turned on. Hence, by delaying timing when the switch SW 1  is turned off for a given time later than a switching between transistors T 1  and T 2 , the switch SW 1  is turned off after voltage levels of the both terminals of the operational amplifier  41  become substantially the same and the voltage is held in the capacitor C 21 . 
     In  FIG. 1 , the coil voltage VL may become a voltage according to an input voltage Vi during the on-period of the transistor T 1 . The coil voltage VL changes according to a coil current IL by the on resistance of the transistor T 2  on the synchronization side during the off period of the transistor T 2 . The coil current IL decreases during the off-period, and the coil voltage VL rises according to the coil current IL (refer to  FIG. 4 ). 
     The operational amplifier  41  outputs a voltage that is obtained by amplifying a potential difference of both terminals as a slope voltage VS. The voltage VS corresponds to a potential difference between the coil voltage VL and the hold voltage VLs. In other words, the voltage VS corresponds to a change amount of the coil current IL (ripple component) during the off-period. As illustrated in  FIG. 4 , the slope voltage VS becomes 0 V (a level indicated by the dotted line) during the on-period, and gradually increases when the switch SW 1  is turned off. 
     As described above, the current gradient detection circuit  27  may detect a ripple component in the coil current IL, and may generate a slope voltage VS according to the detection result. Accordingly, a DC-like increase and decrease in a current at the converter unit  11  by a load (not illustrated) coupled to the output terminal Po may not influence the slope voltage VS. Thus, a stable output voltage Vo may be generated compared with a method that detects an output current. 
     As described above, according to aspects of the embodiment, the following effects may be achieved. 
     (1) The current gradient detection circuit  27  may detect only a ripple component in the coil current IL and may generate a slope voltage VS according to the detected result. The adder circuit  28  may generate a reference voltage VR 1  by adding a slope voltage VS to a standard voltage VR 0 . The comparator  21  compares a feedback voltage VFB according to the output voltage Vo with the reference voltage VR 1 , and may generate a signal VC 1  according to the comparison result. The control circuit  12  turns on and off the transistors T 1  and T 2  of the converter unit  11  based on the signal Vc 1 . Consequently, the output voltage Vo may be stabilized because the transistors T 1  and T 2  may turned on and off periodically even for the output voltage Vo with a small ripple.
 
(2) The current gradient detection circuit  27  may detect only a ripple component in the coil current IL, and may generate a slope voltage VS according to the detected result. Accordingly, a DC like increase and decrease in a current at the converter unit  11  by a load coupled to the output terminal Po may not influence the slope voltage VS. Thus, a stable output voltage Vo may be generated compared with a method that detects an output current.
 
(3) A resistance value of the ESR due to couple the smoothing capacitor C 1  may be made small, thus a laminated ceramic capacitor may be used as the capacitor C 1  and miniaturization and cost reduction of the DC-DC converter may be achieved.
 
     Aspects of the above described embodiment may be implemented by the following modes. 
     In the above described embodiment, the reference voltage VR 1  is generated by adding a slope voltage VS according to a ripple component in the coil current IL to the standard voltage VR 0  and compares the reference voltage VR 1  with a feedback voltage VFB according to the output voltage Vo. In other words, in the output voltage Vo and the standard voltage VRo, the slope voltage is added to the standard voltage VR 0  side. Alternatively, the slope voltage may be added to the output voltage Vo side. 
     For example, as illustrated in  FIG. 6 , a control circuit  12   a  may include an adder circuit  28   a . A first input terminal of the adder circuit  28   a  is coupled between the resistors R 1  and R 2  and to which a feedback voltage VFB is supplied. A second input terminal of the adder circuit  28   a  is coupled to the current gradient detection circuit  27  and the slope voltage VS is supplied. The adder circuit  28   a  adds a voltage that is obtained by inverting a voltage supplied to the second input terminal to the voltage supplied to the first input terminal, in other words, a feedback voltage VF 2  is generated by subtracting the slope voltage VS from the feedback voltage VFB as illustrated in  FIG. 7 . The comparator  21  compares the feedback voltage VF 2  output from the adder circuit  28   a  with the standard voltage VRO (reference voltage) supplied from the standard power supply E 1  and outputs a signal Vc 1  according to the comparison result. The DC-DC converter configured by the above described manner may achieve substantially the same effects as the above descried embodiment. 
     In the above embodiment, the timer circuit  23  is configured so as to output an H level pulse signal S 2  after a given time elapses from the rising timing of the signal S 1  depending on the input voltage Vi and the output voltage Vo. The configuration of the timer circuit  23  may be appropriately changed. 
     For example, the timer circuit  23  may be configured so as to output a signal S 2  after a fixed time elapses. Alternatively, the timer circuit  23  may be configured so as to output the signal S 2  by timing depending only on the input voltage Vi. 
     According to aspects of the above embodiment, MOS transistors are disclosed as examples of the switch circuit and the second switch circuit; however, bipolar transistors may be used. Moreover, a switch circuit that includes a plurality of transistors may be used. 
     According to aspects of the above embodiment, the coil voltage VL changes according to the coil current IL by the on resistance of the transistor T 2  on the synchronization side. Hence, a configuration of the switch on the synchronization side may be appropriately changed if the coil voltage VL may be changed according to the coil current IL. For example, a serial circuit of a resistor and a diode may be coupled instead of the transistor T 2 . 
     According to aspects of the above embodiment, the control signal DH is supplied to the delay circuit  42  illustrated in  FIG. 3 , any signal that corresponds to the on-period or off-period of the transistor T 1  may be supplied, and the signals SH and S 1  illustrated in  FIG. 1  may be supplied to the delay circuit  42 . Furthermore, the signals DH and SH for turning on and off the transistor T 2  may be logically inverted and be supplied to the delay circuit  42 , or the signals DH and SH may be supplied to the delay circuit  42  and the signals may be logically inverted in the delay circuit. 
     All examples and conditional language recited herein are intended for pedagogical purposes to aid the reader in understanding the invention and the concepts contributed by the inventor to furthering the art, and are to be construed as being without limitation to such specifically recited examples and conditions, nor does the organization of such examples in the specification relate to a showing of the superiority and inferiority of the invention. Although aspects of the embodiments) of the present inventions have been described in detail, it should be understood that the various changes, substitutions, and alterations could be made hereto without departing from the spirit and scope of the invention.