Abstract:
Quadrature clocking schemes are widely used in modern communications systems, but often suffer from phase imbalance. Conventional solutions that attempt to address this phase imbalance, however, are generally large and use a substantial amount of power. Here, however, a correction circuit is provided that can locally correct for phase imbalance without the need for bulky and high power consuming circuitry.

Description:
TECHNICAL FIELD 
     The invention relates generally to clocking and, more particularly, to quadrature correction for a clock path. 
     BACKGROUND 
     Quadrature clocking schemes are widely used in modern communications systems, but often suffer from phase imbalance. The initial clock phases are typically generated by a central phase locked loop (PLL), but, during the routing and distribution process, the in-phase (I) and quadrature (Q) phases may become misaligned, particularly if the phases are provided across chip to multiple locations. Quadrature skew can be a phase deviation between the I and Q clocks from the ideal 90°, and quadrature skew is typically caused by the propagation time difference of the I and Q signals transmitting through corresponding channels. Differences in conductor length, trace geometry, via stubs, loading, buffer mismatch, supply differences and some other factors introduce quadrature skew. As the period of the clock decreases, quadrature skew is expected to become more pronounced. Additionally, duty cycle distortion can also be present. Quadrature skew and duty cycle distortion increase random and deterministic jitter and degrade the eye of a receiver. 
     There have been some attempts to solve problems associated with quadrature skew and duty cycle distortion, and, in  FIG. 1 , an example of a conventional clocking scheme or system  100  can be seen. Here, transmitter  102  transmits a clock signal CLK over a transmission medium  104  to a polyphase circuit  106  (which is adjacent to or “close to” a receiver  108 ). The polyphase circuit  106  can then generate the I (PH 1  and PH 3 ) and Q (PH 2  and PH 4 ) signals locally so as to avoid the quadrature skew and duty cycle distortion, which can occur by transmission these signals PH 1  to PH 4  over long transmission lines (i.e.,  104 ). A problem with this system  100 , however, is that the polyphase circuit  106  is generally comprised of PLLs and/or delay locked loops (DLLs) or other circuits, which can occupy a large amount of area and can consume a substantial amount of power. Therefore, there is a need for an improved clocking scheme or system. 
     Some other conventional circuits are: U.S. Pat. No. 6,084,452; U.S. Pat. No. 6,819,728; U.S. Pat. No. 6,933,759; and U.S. Patent Pre-Grant Publ. No. 2002/0085658. 
     SUMMARY 
     A preferred embodiment of the present invention, accordingly, provides an apparatus. The apparatus comprises a plurality of inner correction loops that each receive a pair of input signals and that each perform duty cycle correction for its pair of input signals; and an outer correction loop that is coupled to each of the inner correction loops, wherein the outer correction loops deskews each pair of input signals. 
     In accordance with a preferred embodiment of the present invention, each inner correction loop further comprises: an amplifier that receives its pair of input signals; and a duty cycle correction circuit that is coupled to the amplifier. 
     In accordance with a preferred embodiment of the present invention, the outer correction loop further comprises: an error detector that is coupled to each of the inner correction loops; an error amplifier that is coupled to the error detector; and a steering circuit that is coupled to the error amplifier and to each of the inner control loops. 
     In accordance with a preferred embodiment of the present invention, the error detector further comprises: a logic circuit that is coupled to each of the inner correction loops; a compensator that is coupled to the logic circuit; and a charge pump that is coupled between the compensator and the error amplifier. 
     In accordance with a preferred embodiment of the present invention, an apparatus is provided. The apparatus comprises a first amplifier that receives a first signal and a second signal; a second amplifier that receives a third signal and a fourth signal; a first duty cycle correction circuit that is coupled to the first amplifier; a second duty cycle correction circuit that is coupled to the second amplifier; an error detector that is coupled to the first amplifier and the second amplifier, wherein the error detector detects skew; a error amplifier that is coupled to the error detector; and a steering circuit that is coupled to the error amplifier, the first amplifier, and the second amplifier. 
     In accordance with a preferred embodiment of the present invention, the error detector further comprises: a logic circuit that is coupled to the first and second amplifiers; a compensator that is coupled to the logic circuit; and a charge pump that is coupled between the compensator and the error amplifier. 
     In accordance with a preferred embodiment of the present invention, the logic circuit further comprises: an XOR gate that is coupled between the first amplifier and the compensator; and an XNOR gate that is coupled between the second amplifier and the compensator. 
     In accordance with a preferred embodiment of the present invention, the compensator further comprises: a first inverter that is coupled to the XOR gate; a second inverter that is coupled to the XNOR gate; a third inverter that is coupled between the first inverter and the charge pump; a fourth inverter that is coupled between the second inverter and the charge pump; and a pair of cross-coupled inverters coupled to a node between the first and third inverters and to a node between the second and fourth inverters. 
     In accordance with a preferred embodiment of the present invention, each of the first and second amplifier further comprises: an input stage; an output stage that is coupled to the input stage, at least one of the first and second duty cycle correction circuits, and the error detector; and a steering stage that is coupled to the input stage and the steering circuit. 
     In accordance with a preferred embodiment of the present invention, the input stage further comprises: a pair of bias transistors that are each coupled to a voltage rail; a pair of differential input transistors, wherein each transistor of the pair is coupled to one of the bias transistors and to the steering stage. 
     In accordance with a preferred embodiment of the present invention, the steering stage further comprises a steering transistor that is coupled between the input stage and ground and that is coupled to the steering circuit. 
     In accordance with a preferred embodiment of the present invention, the output stage further comprises a plurality of output stages, and wherein each output stage is coupled to the steering stage. 
     In accordance with a preferred embodiment of the present invention, an apparatus is provided. The apparatus comprises a first amplifier having: a first input stage that receives a first signal and a second signal; a first output stage that is coupled to the first input stage; and a first steering stage that is coupled to the first input stage; a second amplifier having: a second input stage that receives a third signal and a fourth signal; a second output stage that is coupled to the second input stage; and a second steering stage that is coupled to the second input stage; a first duty cycle correction circuit that is coupled to the first output stage of the first amplifier; a second duty cycle correction circuit that is coupled to the second output stage of the second amplifier; an error detector having: a first logic gate that is coupled to the first output stage of the first amplifier; a second logic gate that is coupled to second output stage of the second amplifier; a compensator that is coupled to the first and second logic gates; and a charge pump that is coupled to the compensator; a error amplifier that is coupled to the charge pump of the error detector; and a steering circuit that is coupled to the error amplifier, first steering stage of the first amplifier, and the second steering stage of the second amplifier. 
     In accordance with a preferred embodiment of the present invention, the first logic gate further comprises an XOR gate, and wherein the second logic gate further comprises an XNOR gate, and wherein the compensator further comprises: a first inverter that is coupled to the XOR gate; a second inverter that is coupled to the XNOR gate; a third inverter that is coupled between the first inverter and the charge pump; a fourth inverter that is coupled between the second inverter and the charge pump; and a pair of cross-coupled inverters coupled to a node between the first and third inverters and to a node between the second and fourth inverters. 
     In accordance with a preferred embodiment of the present invention, the steering circuit further comprises: a first PMOS transistor that is coupled to a voltage rail at its source and that receives a bias voltage at its gate; a second PMOS transistor that is coupled to the drain of the first PMOS transistor at its source and that is coupled to the error amplifier at its gate; a third PMOS transistor that is coupled to the drain of the first PMOS transistor at its source and that is coupled to the error amplifier at its gate; a first NMOS transistor that is coupled to the drain of the second PMOS transistor at its drain and gate and that is coupled to ground at its source; and a second NMOS transistor that is coupled to the drain of the third PMOS transistor at its drain and gate and that is coupled to ground at its source. 
     In accordance with a preferred embodiment of the present invention, the bias voltage further comprises a first bias voltage, and wherein each of the first and second output stages and each of the first input stage further comprise: a fourth PMOS transistor that is coupled to the voltage rail at its source and that receives a second bias voltage at its gate; a fifth PMOS transistor that is coupled to the voltage rail at its source and that receives the second bias voltage at its gate; a third NMOS transistor that is coupled to the drain of the fourth PMOS transistor at its drain and at least one of the first and second steering circuits at its source and that receives at least one of the first and third signals at its gate; a fourth NMOS transistor that is coupled to the drain of the fifth PMOS transistor at its drain and at least one of the first and second steering circuits at its source and that receives the second signal at its gate; a sixth PMOS transistor that is coupled to the voltage rail at its source and the drain of the first PMOS transistor at its gate; a seventh PMOS transistor that is coupled to the voltage rail at its source and the drain of the second PMOS transistor at its gate; a first current minor that is coupled between the drains of the sixth and seventh PMOS transistors and at least one of the first and second steering circuits; a fifth NMOS transistor that is coupled to the drain of the sixth PMOS transistor at its drain, at least one of the first and second duty cycle correction circuits at its gate, and at least one of the first and second steering circuits; an eighth PMOS transistor that is coupled to the voltage rail at its source and the drain of the first PMOS transistor at its gate; a ninth PMOS transistor that is coupled to the voltage rail at its source and the drain of the second PMOS transistor at its gate; a second current mirror that is coupled between the drains of the eighth and ninth PMOS transistors and at least one of the first and second steering circuits; and a sixth NMOS transistor that is coupled to the drain of the ninth PMOS transistor at its drain, at least one of the first and second duty cycle correction circuits at its gate, and at least one of the first and second steering circuits at its source. 
     In accordance with a preferred embodiment of the present invention, each of the first and second steering circuits further comprises: a seventh NMOS transistor that receives a third bias voltage at its gate and that is coupled to ground at its source; and an eighth NMOS transistor that is coupled to the gate of at least one of the first and second NMOS transistors at its gate, the drain of the seventh NMOS transistor at its drain, and ground at its source. 
     In accordance with a preferred embodiment of the present invention, the bias voltage further comprises a first bias voltage, and wherein each of the first and second output stages and each of the first input stage further comprise: a fourth PMOS transistor that is coupled to the voltage rail at its source and that receives a second bias voltage at its gate; a fifth PMOS transistor that is coupled to the voltage rail at its source and that receives the second bias voltage at its gate; a third NMOS transistor that is coupled to the drain of the fourth PMOS transistor at its drain and at least one of the first and second steering circuits at its source and that receives at least one of the first and third signals at its gate; a fourth NMOS transistor that is coupled to the drain of the fifth PMOS transistor at its drain and at least one of the first and second steering circuits at its source and that receives the second signal at its gate; a first current mirror that is coupled to the voltage rail; a second current mirror that is coupled to the voltage rail; a third current mirror that is coupled to the first current minor, and the drain of the fifth PMOS transistor, and ground; a fourth current minor that is coupled to the second current mirror, and the drain of the fourth PMOS transistor, and ground; a third NMOS transistor that is coupled to the first current minor at its drain and at least one of the first and second duty cycle correction circuits at its gate; a fourth NMOS transistor that is coupled to the second current mirror at its drain and at least one of the first and second duty cycle correction circuits at its gate; and a fifth NMOS transistor that receives a second bias voltage at its gate, that is coupled to the sources of the third and fourth NMOS transistors at its drain, and that is coupled to ground at its source. 
     In accordance with a preferred embodiment of the present invention, each of the first and second steering circuits further comprises: a sixth NMOS transistor that receives a third bias voltage at its gate and that is coupled to ground at its source; and a seventh NMOS transistor that is coupled to the gate of at least one of the first and second NMOS transistors at its gate, the drain of the seventh NMOS transistor at its drain, and ground at its source. 
     The foregoing has outlined rather broadly the features and technical advantages of the present invention in order that the detailed description of the invention that follows may be better understood. Additional features and advantages of the invention will be described hereinafter which form the subject of the claims of the invention. It should be appreciated by those skilled in the art that the conception and the specific embodiment disclosed may be readily utilized as a basis for modifying or designing other structures for carrying out the same purposes of the present invention. It should also be realized by those skilled in the art that such equivalent constructions do not depart from the spirit and scope of the invention as set forth in the appended claims. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which: 
         FIG. 1  is a circuit diagram of an example of a conventional system; 
         FIG. 2  is a circuit diagram of an example of a system in accordance with a preferred embodiment of the present invention; 
         FIG. 3  is a circuit diagram of an example of the correction circuit of  FIG. 2 ; 
         FIG. 4  is a circuit diagram of an example of the quadrature error detector of  FIG. 3 ; 
         FIG. 5  is a circuit diagram of an example of the steering circuit of  FIG. 3 ; and 
         FIGS. 6 and 7  are circuit diagrams of examples of the amplifiers of  FIG. 3 . 
     
    
    
     DETAILED DESCRIPTION 
     Refer now to the drawings wherein depicted elements are, for the sake of clarity, not necessarily shown to scale and wherein like or similar elements are designated by the same reference numeral through the several views. 
     Turning to  FIG. 2 , an example of a system  200  in accordance with a preferred embodiment of the present invention can be seen. System  200  is different from system  100  in that transmitter  202  provides the I channel (clocks signals CLK 1  and CLK 3 ) and Q channel (clock signals CLK 2  and CLK 4 ) over transmission medium  204  (which can be up to several millimeters in length). A correction circuit  206  receives these clock signals CLK 1  to CLK 4  and corrects for quadrature skew and duty cycle distortion to locally generate signals PH 1  to PH 4  for receiver  108 . Additionally, clock signals CLK 1  to CLK 4  generally correspond to 0°, 90°, 180°, and 270° phases of a master or initial clock signal (i.e., clock signal CLK from  FIG. 1 ). 
     In  FIG. 3 , an example of the correction circuit  206  can be seen in greater detail. As shown, the correction circuit  206  generally comprises amplifiers  302  and  303 , duty cycle correction circuits (or DCC)  304 - 1  and  304 - 2 , quadrature error detector  308 , error amplifier  306 , steering circuit  312 , and buffer  310  that are arranged into a set of inner loops and an outer loop. The inner loops generally correct for duty cycle distortion by using feedback from the duty cycle correction circuits  304 - 1  and  304 - 2 , and the output loop, on the other hand, generally corrects for quadrature skew by using feedback from the quadrature error detector  308 , error amplifier  306  and steering circuit  312 . Additionally, the inner loops typically settle faster than the outer loop. 
     Looking first to the inner or duty cycle correction loop, “opposite” clock phases CLK 1 /CLK 3  or CLK 2 /CLK 4  (which generally correspond to differential I and Q channels, respectively) are used. As shown in  FIG. 3 , each of amplifiers  302  and  303  receives one of the pairs of “opposite” phase clock signals CLK 1 /CLK 3  or CLK 2 /CLK 4 . A reason for using the “opposite” phases is that any overlaps in the “phases” can indicate an error in duty cycle. The duty cycle correction circuits  304 - 1  and  304 - 2  can then generate a correction feedback for amplifiers  302  and  303 , respectively, based on any overlap. For example, if there is an overlap between the “ON” times of clock signal CLK 1  (0° phase) and clock signal CLK 3  (180° phase), the duty cycle of one of these clock signals CLK 1  or CLK 3  may be too long, which can be corrected through feedback to amplifier  302 . These inner loops can then converge when signals PH 1  and PH 3  (for the I channel) and signals PH 2  and PH 4  (for Q channel) have about 50% duty cycle. 
     The outer loop operates on a different principal. For the phase difference ( Δ Φ in ) between the I and Q channels and the phase difference ( Δ Φ out ) of the output clocks signals PH 1 /PH 2  and PH 3 /PH 4 , the following control equation can be used to model the outer loop:
 
( Δ Ψ in −π/2)−( Δ Ψ out −π/2) H ( s ) A* 2 πFΔt=   Δ Ψ out ,  (1)
 
where F is the clock frequency, Δt is the delay stage coefficient, A is loop gain, and H(s) is the transfer function. By collecting terms for equation (1), the following equation can be obtained:
 
                     Ψ   out           Δ         =           Ψ   in           Δ         -     π   /   2         1   +       H   ⁡     (   s   )       ⁢   A   *   2   ⁢   π   ⁢           ⁢   F   ⁢           ⁢   Δ   ⁢           ⁢   t         +       π   2     ⁢       H   ⁢     (   s   )     ⁢   A   *   2   ⁢   π   ⁢           ⁢   F   ⁢           ⁢   Δ   ⁢           ⁢   t       1   +       H   ⁡     (   s   )       ⁢   A   *   2   ⁢   π   ⁢           ⁢   F   ⁢           ⁢   Δ   ⁢           ⁢   t                     (   2   )               
As can be seen in equation (2), the initial input quadrature error can be suppressed by the loop gain (A), which can also maintain the phase difference of the output I/Q clock at π/2. The outer loop can basically be a high pass system and can be implemented as a first order loop. Higher order loops can also be implemented to reduce control voltage ripple by including some additional resistor-capacitor (RC) networks to the charge pump  408  (shown below) or adding one more gain stages with an RC network after the quadrature error detector  308 . The latter will allow some more freedom to control gain and level shifting.
 
     Now, turning to  FIG. 4 , an example of the quadrature error detector  308  (which is part of the outer loop) can be seen in greater detail. In contrast to the inner loops, the output loop, and more particularly, the quadrature error detector  308  compares adjacent phase clock signals PH 1 /PH 2  and PH 3 /PH 4 . As shown, the quadrature error detector  308  generally comprises a logic circuit (i.e., XOR gate  402  and XNOR gate  403 ), a compensator  404  (i.e., inverters  406 - 1  to  406 - 6 ), and a charge pump  408  (i.e., PMOS transistors Q 1  through Q 4 , NMOS transistors Q 3  through Q 7 , capacitors C 1  and C 2 , resistors R 1  and R 2 , and bias voltage BIAS 1  received at the gate of transistor Q 7 ). In operation, XOR gate  403  and XNOR gate  403  convert a skew error into duty cycle errors. Typically, the outputs of XOR gate  403  and XNOR gate  403  are each a complementary square waves having twice the input frequency with a duty cycle that depends on the relative phase between the I and Q inputs. The resultant duty cycle should vary between 0% and 100%, being 50% if I and Q phases are in quadrature (phase difference 90°), and the differential input to the charge pump  408  should have average value that varies as a function of the relative phase, being 0 when the I/Q phase is at 90°. The charge pump  408  generally integrates and filters the differential input from the logic circuit (i.e., XOR gate  402  and XNOR gate  403 ) so as to convert the differential input into a differential control voltage signal (for error amplifier  306 ) based on the input duty cycle. For charge pump  408 , when duty cycle is 50%, the charging and discharging periods are generally equal so that little to no charge will accumulate across capacitor C 2 . When the duty cycle differs from 50%, the charging and discharging periods are generally imbalanced, charging capacitor C 2 . The series resistors R 1  and R 2  and parallel capacitor C 1  serve to further adjust the frequency properties of the charge pump for accuracy and loop stability inside the outer loop. The compensator  404  generally includes a pair of cross-coupled inverter  406 - 3  and  406 - 4  (which are coupled between inverters  406 - 2 / 406 - 5  and  406 - 1  and  406 - 6 ) to generally eliminate skew error introduced by the mismatch between XOR and XNOR gates  402  and  403  so as to increases the accuracy of the quadrature skew detecting. 
     Turning to  FIG. 5 , an example of the steering circuit  312  can be seen in greater detail. The steering circuit  312  generally comprises a bias transistor Q 39  (which can be a PMOS transistor that is coupled to a voltage rail (i.e., VCC) and which and which receives a bias voltage BIAS 2 ), a differential input pair Q 8  and Q 9  (which can be PMOS transistors), and diode-connected transistors Q 10  and Q 11  (which can be NMOS transistors). The differential input pair Q 8  and Q 9  receive signals IADJ and QADJ signals, which are issued from charge pump  408  and amplified by error amplifier  306 , and the resulting currents (from transistors Q 8  and Q 9 ) control the tail currents in amplifiers  302  and  303  through diode-connected transistors Q 10  and Q 11 . 
     In  FIG. 6 , an example of amplifier  302  or  303  (labeled  302 - 1  or  303 - 1 ) can be seen in greater detail. As shown, amplifier  302 - 1  or  303 - 1  generally comprises an input stage  602 , a steering stage  606 , and output stages  604 - 1  and  604 - 2  (which are typically differential delay elements). For input stage  602 , differential input transistors Q 18  and Q 19  (which can be NMOS transistors) are cascoded with bias transistors Q 16  and Q 17  (which receive bias voltage BIAS 3  and which can be PMOS transistors). Each of output stages  604 - 1  and  604 - 2  generally comprises differential input transistors Q 12 /Q 13  and Q 20 /Q 21  (that are each generally coupled to the input stage  602 ), current minors Q 14 /Q 15  and Q 22 /Q 23  (which are each generally coupled to the respective differential input transistors Q 12 /Q 13  or Q 20 /Q 21 ), and a control transistors Q 24  and Q 27  (which receive control signals from the respective duty cycle control circuit  304 - 1  or  304 - 2  to vary the delay of the output stages  604 - 1  or  604 - 2  accordingly). Typically, the duty cycle correction circuits  304 - 1  and  304 - 2  measure duty cycle distortion of signals PH 1  to PH 4  and correct the distortion by changing the switching threshold of the output stages  604 - 1  and  604 - 2  for each of amplifiers  302 - 1  and  303 - 1 . The steering circuit  606  generally comprises a bias transistor Q 25  (which receives a bias voltage BIAS 4  and which can be an NMOS transistor) and a transistor Q 26  (which can be an NMOS transistor and collectively with transistor Q 10  or Q 11  can form a current mirror to control the tail current in amplifier  302  or  303 ). The change in bias current (through steering stage  606 ) can affects the common mode output and, in turn, changes the relative delay of the I and Q paths, speeding up one while slowing down the other. The delay input stages (i.e.,  604 - 1  and  604 - 2 ) are typically biased at nominal currents to generally prevent the outer loop from saturating. Additionally, using separate physical paths for the inner and outer loops generally insures the two control mechanisms work with minimal interaction. 
     Turning to  FIG. 7 , an alternative arrangement for amplifier  302  or  303  (labeled  302 - 2  or  303 - 2 ) can be seen in greater detail. Here, the difference between amplifier  302 - 1  or  303 - 1  and amplifier  302 - 2  or  302 - 3  lies in the output stage  702 . This output stage  702  is not generally controlled by steering stage  606  (as the output stages  604 - 1  and  604 - 2  are in amplifier  302 - 1  or  303 - 1 ). Output stage  702  generally comprises differential input transistors Q 30  and Q 31  (which receive signals from duty cycle correction circuit  304 - 1  or  304 - 2  and which may be NMOS transistors), a bias transistor Q 38  (which receives bias voltage BIAS 5  and which can be an NMOS transistor), and current minors Q 28 / 29 , Q 32 /Q 33 , Q 34 /Q 35 , and Q 36 /Q 37 . In operation, duty cycle error is detected by an integrator in the duty cycle correction circuit  304 - 1  or  304 - 2 , and the error is corrected by DC offset adjustment voltages applied to transistors Q 30  and Q 31 . Quadrature error is corrected by adjusting the tail current in steering stage  606  to accelerate or decelerate the slew rate. Additionally, bias voltage BIAS 5  can provide a residue current for during start up. 
     Having thus described the present invention by reference to certain of its preferred embodiments, it is noted that the embodiments disclosed are illustrative rather than limiting in nature and that a wide range of variations, modifications, changes, and substitutions are contemplated in the foregoing disclosure and, in some instances, some features of the present invention may be employed without a corresponding use of the other features. Accordingly, it is appropriate that the appended claims be construed broadly and in a manner consistent with the scope of the invention.