Abstract:
A charge pump circuit includes a pair of series switching devices coupled between an output node of the circuit and an input node. A power stage drives a charge transfer capacitor which is coupled to an intermediate node between the series switching devices. The power stage has an input coupled to the input node of the circuit, The power stage further includes a bootstrap capacitor for maintaining a conductive state during an entire half of a cycle of a period of oscillation of a local oscillator. The series switching devices may be driven in phase opposition by either a CMOS invertor or a pair of comparators.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention concerns a charge pump circuit for implementing a voltage multiplier in integrated circuits operating with a relatively low supply voltage. The circuit is particularly suited for driving a high-side driver power device of a half-bridge output stage employing integrated DMOS transistors. 
     2. Discussion of the Related Art 
     Integration in the same chip of complex systems often composed of distinct circuit sections that are necessarily realized with structurally and functionally different devices, erasing and programming of read only memory cells, optimization of the driving conditions of integrated power devices in a half-bridge configuration, require or benefit from the availability of a biasing or driving voltage higher than the supply voltage of the integrated circuit. Often these requirements are satisfied by integrating special voltage multiplying circuits capable of producing, on an output ballast capacitance (storage capacitance) a voltage that is higher than the supply voltage (VCC). These circuits are commonly referred to as voltage multipliers or charge pump circuits. 
     A basic diagram of a charge pump circuit is depicted in FIG. 1. 
     Basically, the capacitor C2 stores electric charge that is fed thereto by transferring (pumping) the charge momentarily stored in a transfer capacitance C1 through a transfer diode D2. The intermittent charging of C1 through a diode D1 connected to VCC, and the discharging of the electric charge stored therein into the output capacitance C2, are controlled by a free-running local oscillator (OSC LDC)10. At steady state, the circuit operates as follows. 
     During an initial half cycle, the output of the local oscillator 10 is low and the capacitance C1 charges through the diode D1 to a voltage given by: VCC-Vd, where Vd represents the voltage drop across the charge diode D1. In the following half cycle, the output of the oscillator goes to a voltage VR, the node A assumes the voltage given by: VCC+VR-Vd, and the capacitance C1, discharges the electric charge that was stored therein during the preceding half cycle in the output storage capacitance C2. The capacitance C2, assumes the voltage VCP given by the following relation: 
     
         VCP=VCC+VR-2.Vd 
    
     It may be observed that for: VR=VCC, the circuit becomes substantially a voltage duplicator and provides an output voltage equal to twice the supply voltage less the voltage drop across the diodes. 
     Of course, the circuit may be composed of N basic circuit modules, in order to obtain an output voltage that is N-times the supply voltage less N-times the ohmic drop across a diode. 
     A practical embodiment of this type of circuit is depicted in FIG. 2. A regulated reference voltage VR is obtained, in the most simple case, by using a common voltage regulating stage composed of the transistor Q1, the resistor R1, the diode D3 and the Zener diode DZ1. By considering the voltage across the direct biased base-emitter junction of transistor Q1, given approximately by: Vbel=Vd3, where Vd3 is the voltage across the diode D3, the regulated voltage VR will be given by the following relations: ##EQU1## where VCC is the supply voltage, VZ is the Zener voltage and Vdrop is the sum of the junction voltage Vbel and of the voltage drop across the resistance R1 due to the current absorbed by the base of the regulating transistor Q1. 
     In a typical circuit as depicted in FIG. 2, Vdrop is about 1.2 V. However in a large number of applications, a relatively high current may be required and therefore the transistor Q1 is often replaced by a Darlington pair. In this last instance, Vdrop is about 2 V. 
     The voltage regulating stage provides a regulated supply to the local oscillator 10 and to the power stage that is made up of the two MOS transistors M1 and M2. The power stage has the function of driving the charge and discharge of the pump capacitance C1. When the load connected to an output (VCP) of a multiplied voltage source is relatively heavy and/or the transistors that are connected to the VCP line switch at a high frequency, relatively large capacitances are needed both for C1 (on the order of 10 nF) and for C2 (several hundreds of nF). As a consequence, the power stage (M1-M2) must be capable of delivering relatively large peaks of current (in the order of ten mA). The drawbacks of a conventional circuit configuration as the one described above are the following: 
     a) throughout the range of the supply voltage, the voltage drop across the transfer diodes D1 and D2 (equivalent to about 1.4 V) is &#34;subtracted&#34; from the output voltage VCP, as may be observed by considering the following relation: ##EQU2## b) For a relatively low supply voltage (e.g. VCC&lt;VZ+Vdrop), the voltage drop across the voltage regulator (Vdrop=2 V) subtracts from the available output voltage VCP, as may be observed from the following relation: 
     VCP=2*VCC-Vdrop-2Vd for VCC&lt;VZ+Vdrop; Both voltage drops contribute to limit the efficiency of the circuit when operating with a low supply voltage; 
     c) The current driving the pump capacitance C1, which may have relatively large peaks is drawn from the regulated voltage line VR. This implies the need of properly designing the voltage regulator, to that is, employing large size components. 
     SUMMARY OF THE INVENTION 
     An objective of the present invention is to provide a charge pump circuit that will maintain its efficiency with a low supply voltage and that will not require a high-current voltage regulating device. 
     This objective is accomplished by the circuit of the present invention. The circuit includes electric charge transfer devices that are switching elements which replace the diodes according to the prior art. As a result, the voltage drops across the charge transfer devices is substantially reduced. Moreover, by employing a half-bridge power stage, provided with a &#34;bootstrap&#34; capacitance, the voltage drop (Vdrop) across the voltage regulating stage is avoided. 
     In one embodiment, the two switches that replace the two charge transfer diodes to and from the pump capacitance C1, include a first series pair of complementary transistors, connected in phase opposition, and driven by a second CMOS pair of transistors (comprising an inverter). The state of conduction and of interdiction of the two complementary transistors forming the first CMOS pair of transistors is maintained for an entire half cycle of oscillation by the second CMOS pair (inverter) which, together with the first CMOS pair, constitute a bistable circuit, commonly referred to as a latch or flip-flop. 
     The half-bridge power stage is functionally connected between the supply node VCC and ground and may be formed with DMOS devices (e.g. with VDMOS transistors). The power device constituting the so-called the high-side driver of the half-bridge power stage is controlled by an inverting stage. The inverting stage is driven by the signal generated by the local oscillator and is functionally connected between the regulated voltage line and ground. Conversely, the other power device of the power stage (low-side driver) may be driven directly by the signal produced by the local oscillator. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The different aspects and advantages of the circuit of the invention will become more evident through the following description of several important embodiments and by referring to the attached drawings, wherein 
     FIGS. 1 and 2 show a charge pump circuit according to the prior art, as described above; 
     FIG. 3 shows a pump circuit made in accordance with the present invention; 
     FIG. 4 is a partial schematic diagram of an alternative embodiment of the charge transfer devices; 
     FIG. 5 is a block diagram of a voltage multiplier including an even number of the charge pump circuits of the present invention; and 
     FIG. 6 is a block diagram of a voltage multiplier including an odd number of the charge pump circuits of the present invention. 
    
    
     DETAILED DESCRIPTION 
     By referring to the diagram of FIG. 3, the operation of the charge pump stage of the invention is as follows. 
     1. OPERATION OF THE POWER STAGE OF THE CHARGE PUMP CIRCUIT 
     During a first half cycle, when the output of the local oscillator 10 is high (at the regulated voltage VR), the power transistor M1 (high-side driver) is OFF and the power transistor M2 (low-side driver) is ON. Therefore, the output node M of the half-bridge power stage is at ground potential and the bootstrap capacitance Cb charges through the transistor Q2 to the regulated voltage VR present on the node R. 
     During the successive half cycle of oscillation, when the output of the oscillator commutes from the voltage VR to ground potential, M2 is switched-OFF and M1 is switched-ON, thus &#34;shortcircuiting&#34; the node R (to which the bootstrap capacitance Cb is connected) to the gate of M1, through the transistor M3. As a consequence, the output node M of the half-bridge stage takes a voltage VR&#39; and the power transistor M1 is kept on during the entire half cycle by the charge stored in the bootstrap capacitance Cb. The voltage VR&#39;, assumed by the driving node M, will have the following values: 
     VR&#39;=VZ+VBE for VCC&gt;VZ+VBE 
     VR&#39;=VCC for VCC&lt;VZ+VBE 
     The transistor Q3 and the Zener DZ2 have the purpose of limiting the voltage on the node M to the value VR&#39;, given by the first one of the above-indicated equations. In fact, once the required voltage is reached (i.e. VZ+VBE), the transistor Q3 switches-OFF the power transistor M1, thus preventing an unnecessary consumption of current through the power transistor M1 itself. 
     2. Operation of the Charge Transfer Switches 
     Once the initial switch-ON transient period has terminated, the output charge storing capacitance C2 assumes a potential given by VCC-2.Vd&#39;, where Vd&#39; represents the voltage drop across the &#34;body&#34; diode of a generic power MOS structure (if M7 and M8 are integrated in the same pocket, the starting voltage assumed by the capacitance C2 will be given by VCC-Vd&#34;, where Vd&#34; represents the voltage drop across the &#34;body&#34; diode of the pocket). 
     As long as the voltage VCP-VCC remains lower than the threshold voltage of the CMOS transistors (M5, M6, M7 and MS), C2 will continue to charge through the diodes intrinsic to the integrated structure of the transistors M5 and M6. As soon as the voltage VCP-VCC becomes greater than the threshold voltage of the CMOS transistors, M5 and M6, driven by the inverting stage constituted by the CMOS pair M7 and MS, will conduct alternately, cooperating to &#34;pump&#34; electric charge in C2. 
     The steady state operation may be described as follows. 
     During the half cycle of oscillation when the driving node M is at ground potential, the capacitance C1 charges to VCC through the MOS transistor M5, which is kept ON by the inverting stage formed by the pair of CMOS transistors M7 and M8. In fact, the transistors M5, M6, M7 and M8 are configured in a positive reaction mode (i.e. in a configuration that implements substantially a latch or a flip-flop), and as a consequence, the condition M5on, M6on, persists until the next switching of the driving node M. 
     During the following half cycle, when the driving node M assumes the VR&#39; potential, the electric charge that was stored in the transfer capacitance C1 during the preceding half cycle transfers thus charging it, into the output storage capacitance C2 through the transistor M6, which is maintained conducting during this half cycle and until a successive switching of the driving node M because of the positive reaction that is implemented through M7-M8. 
     From the above considerations, it may be verified easily that the output voltage VCP assumes the following values: 
     VCP=VCC+VZ+VBE for VCC&gt;VZ+VBE 
     VCP=2*VCC for VCC&lt;VZ+VBE 
     The above formulas are valid for purely capacitive loads. 
     Of course, when the current absorbed by the load has nonnegligible value, the internal ON-resistance of transistors M1, M2 and M5 and M6, must be taken into consideration. 
     The circuit of the invention has remarkable advantages as compared to the known circuits. The main advantages may be indicated as follows: 
     a) reduction of the maximum (multiplied) output voltage by a quantity equivalent to the voltage drop across the transfer diodes of the known circuits can be avoided throughout the range of the supply voltage; 
     b) reduction of the maximum output voltage by a quantity equivalent to the voltage drop (Vdrop) through the voltage regulator, under conditions of low supply voltage, may be avoided too; 
     c) because the half-bridge, power stage made with the power transistors M1 and M2 is electrically connected to the supply line VCC, the high current peaks that occur for driving the transfer capacitance C1 no longer need to be drawn from the voltage regulator circuit. 
     According to an alternative embodiment of the circuit of the invention, depicted in FIG. 4, the efficiency of the circuit may be increased further. 
     As schematically shown in the partial diagram of FIG. 4, the MOS transistors M5 and M6 that functionally replace the diodes D1 and D2 of the conventional circuits, as depicted in FIGS. 1 and 2, may be separately driven in order to avoid cross-currents and return currents at switchings. In practice this may implemented by employing two comparators: A1 and A2, or more simply two inverters, one having a low threshold for driving the N-channel transistor M5 and one having a high threshold for driving the P-channel transistor M6. 
     In case of integration of the charge pump circuit of the invention, it is recommended to place the transistors M5 and M6 into separate pockets and to bias the pocket of M5 to the bulk potential. Should M5 and M6 be in the same tub, or pocket, a parasitic bipolar transistor would discharge C2 in the half period while C1 is being charged, thus making it impossible to reach a steady state condition. Moreover, it is advisable to minimize the gain of the parasitic PNP transistor of the integrated structure of M6, as well as to minimize the parasitic capacitances of the R and S nodes of the circuit depicted in FIG. 3. 
     The charge pump circuit of the invention, illustrated above for a voltage &#34;duplicator&#34;, may constitute a building block for realizing an N-order voltage multiplier, where N may be theoretically as large as desired. 
     An N-even multiplier is depicted in FIG. 5, wherein each three-terminal block (Blk) may be constituted by a circuit such as the one composed of transistors M5, M6, M7 and M8 of FIG. 3, or by a circuit similar to the one composed of transistors M5 and M6 and of the respective driving threshold comparators A1 and A2 of FIG. 4. 
     An N-odd multiplier is depicted in FIG. 6, wherein each three-terminal block (Blk) may be a circuit such as the one composed of transistors M5, M6, M7 and M8 of FIG. 3, or a circuit such as the one composed of transistors M5 and M6 and of the respective driving threshold comparators A1 and A2 of FIG. 4. 
     Having thus described one particular embodiment of the invention, various alterations, modifications, and improvements will readily occur to those skilled in the art. Such alterations, modifications, and improvements are intended to be part of this disclosure, and are intended to be within the spirit and scope of the invention. Accordingly, the foregoing description is by way of example only and is limited only as defined in the following claims and the equivalents thereto.