Abstract:
A method is provided for detecting the skew between parallel light signals generated from a serial data stream. The method can be used with polarization multiplexed signal, as well as with wavelength division multiplexed signals, spatial division multiplexed signals, phase modulated signals, or intensity modulated signals. The method can be used with direct detection schemes as well as with coherent detection schemes. The method is provided with: imprinting dips between a fixed number of transmitted symbols of the parallel signals; detecting an electrical signal related to the dips for each parallel signal; and comparing the electrical signals in delay.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
     This application is a National Stage of International Application No. PCT/JP2009/058036 filed on Apr. 16, 2009, the contents of all of which are incorporated herein by reference in their entirety. 
     TECHNICAL FIELD 
     The present invention relates generally to an optical communication technique, and more particularly to parallel communications, in which one signal at a high bit rate is split and transmitted through parallel channels at lower bit rates, and is reconfigured from the transmitted parallel channels. 
     BACKGROUND ART 
     Through Serial to Parallel conversion, a serial high bit rate channel is split into lower bit rate channels, in order to transmit them as parallel channels between a same transmission start point to a same transmission stop point, while the parallel channels do not have any crosstalk ideally. The parallel channels are then serialized into a high bit rate channel, which contains the transmitted information of the original serial channel. This makes it possible to decrease the effect of distortions and impairments appearing during the transmission, whose effects increase with the bit rate. It also makes it possible to reduce the spectrum width of transmitted signals because the parallel channels have a lower bit rate than the serial channel. 
     Known ways to implement parallel transmissions include: Polarization Multiplexing (PM), in which two signals are transmitted through the same medium by two signals having orthogonal polarizations; Wavelength Division Multiplexing (WDM), in which the parallel channels are transmitted through the same medium by different optical carriers having different wavelengths; Space Division Multiplexing (SDM), in which the parallel channels are transmitted through different mediums, which can be bounded such as a fiber ribbon; combinations of the previous multiplexing formats. 
     PM is recognized as an efficient way to double a transmitted data rate of an optical transmissions system as it makes it possible to carry two light signals on orthogonal polarizations of an optical fiber used as transmission medium. Both of the signals ideally do not interfere as the polarizations are orthogonal. Therefore, it is a way to improve the efficiency of the use of bandwidth of the optical fiber. As the bit rate of a total transmitted signal is doubled, whereas the baud rate of each polarization signal is unchanged, the width of the optical spectrum is unchanged. Therefore, PM does not degrade a tolerance to optical filtering occurring on the path of the light signal transmitted inside a network when compared with one single polarization, although a double of information is transmitted. However, polarization demultiplexing is required in order to recover each of the polarization multiplexed signals. One example of PM is given in “Precise Analysis of Transmission Impairments of Pol-Mux 110 Gb/s RZ-DQPSK with Automatic Pol-Dmux using Straight 2,000-km SMF Line” (ECOC 2008, paper We.1.E.6) by T. Ito and al. 
     One known method to implement PM is to provide two transmitters emitting the same wavelength and then to rotate the polarization of the signal emitted from one transmitter before combining it with the signal emitted from the other transmitter, so that both polarization are orthogonal to each other. Typically, the optical transmitters used in optical transmission systems are provided with a laser source emitting a continuous wave optical carrier signal and an optical modulator. The light carrier signal emitted from a laser source is linearly polarized; the laser source and the optical modulator are connected by a polarization maintaining fiber. Therefore, the light signal emitted from the transmitter is linearly polarized. In this configuration, rotating the polarization in one transmitter can be performed by a polarization controller or simply by rotating one axis of the polarization maintaining fiber carrying the light signal. A light signal resulting from a combination of the light signals emitted from two transmitters in this configuration is polarization multiplexed. 
     An alternative implementation method consists of using a same light source for both of the polarizations. A polarization maintaining coupler can be used to split a light carrier signal emitted from the laser source and each of the signals split by the coupler can be fed to an optical modulator. The polarization rotating scheme and combining scheme are identical to the above-mentioned implementation. 
     A binary data stream to be transmitted with the light signal may be pre-coded and de-serialized into tributary binary data streams; each of the tributary streams may be allocated to one driver, which generates a voltage used to drive an optical modulator imprinting the information to the light signal of one of the multiplexed polarizations. The multiplexed polarizations may be randomly and dynamically rotated while the light signal is transmitted through the fiber, although they remain orthogonal. Therefore, a dynamic polarization de-multiplexing scheme is useful on the receiver side. 
     One implementation of polarization demultiplexing can be qualified of optical polarization demultiplexing. It consists of rotating both of the multiplexed polarizations by a polarization controller so that they can be split by a polarization beam splitter or another polarization separating device. The polarization controller can be actively controlled so that the beam splitter correctly separate the polarization multiplexed signals, even when the polarizations are rotated while being transmitted through the optical fiber. Each of the separated polarizations is received by a separate optical receiver, which decodes the data of the light signal and converts it into a binary data stream. Electrical signals resulting from both of receivers are then serialized and the resultant binary data stream may be decoded. This is described in “Comparison of 100 Gb/s transmission performances between RZ-DQPSK and polarization multiplexed NRZ/RZ-DPSK with automatic polarization de-multiplexer” (OFC 2008, paper JThA46) by T. Ito and al. 
     An alternative method of performing polarization demultiplexing can be qualified of digital polarization demultiplexing. A light signal is received by a coherent receiver. The resultant lightwave signals can be converted into digital signals by analog to digital converters (ADCs). The resultant digital data can be calculated in accordance with appropriate algorithms that can retrieve and separate the data imprinted on each multiplexed polarization. The retrieved data streams are then decoded and serialized. One example is given in “PLL-Free Synchronous QPSK Polarization Multiplex/Diversity Receiver Concept with Digital I&amp;Q Baseband Processing” (Photonics Technology Letters, Vol. 17, No. 4, April 2005) by R. Noe. 
     The binary data stream before serial-to-parallel conversion, emission and transmission through polarization multiplexed light signals and the binary data stream after polarization demultiplexing, reception and parallel-to-serial conversion are ideally identical. Differences between the streams are transmission errors. One data bit will be de-serialized and then allocated to the optical modulator imprinting data on the light signal of one polarization. This data will travel electrically from the de-serializer to the optical modulator, and then it will be transmitted optically from the modulator to an optical polarization combiner. This bit will be transmitted through a medium and received by a receiver. Another data bit of the serial data stream will be de-serialized in the same way but it will be allocated to the optical modulator imprinting data on the light signal of the other polarization. This other bit of data will travel electrically from the de-serializer to the other optical modulator, and then it will be transmitted optically from the other modulator to the optical polarization combiner. This other bit will be transmitted through the medium and received by a receiver. Therefore, both of the considered bits will travel on different electrical paths from the de-serializer to their optical modulators and on different optical paths from their modulators to the polarization combiner. 
     Differences in electrical or optical path lengths for the considered bits results in time difference for the transmission on the multiplexed polarizations. This problem is known as intra-channel skew. If the skew is more than half a symbol period of the signal transmitted on one polarization, the bits travelling through the longer path will arrive to the serializer in the receiver with a delay of more than one bit when compared to the bits travelling through the shorter path. This situation results in a change of the bit sequence during the transmission, i.e. in a transmission error. 
     Assuming 100 Gbit/s Dual Polarization Quadrature Phase Shift Keying (QPSK) transmission, the bit rate of the serial binary data stream is 100 Gbit/s but the symbol rate of the transmitted light signal is 25 Gbaud. Therefore, the symbol period of the light signal on one polarization is 40 ps. This corresponds roughly to the length tolerance of 4 mm of optical fiber for the optical path or 4 mm of wire for the electrical path. Considering the fact that the optical components for each polarization paths are usually connected by spliced fiber, such a tolerance may be hard to achieve on production. Moreover, Differential Group Delay (DGD) accumulated on the fiber during transmission may delay one polarization compared to the other one, and therefore randomly and dynamically change the intra-channel skew. 
     In addition, dynamic network reconfiguration or switching may cause a value of a link skew to change dynamically. If the skew is more than half a symbol period, the transmission is not error free with respect to the received data after parallel-to-serial conversion. Therefore, there is a need for monitoring the delay between polarization multiplexed signals in order to use this information to correct this delay and achieve correct transmission of data. 
     Various approaches have been proposed in order to solve this problem. The most obvious one is to use the Forward Error Correction (FEC) data of the serialized received data. This enables to check if the data is correct, in which case the delay between polarizations does not need to be corrected. If the error rate is more than 0.5 or in the order of 0.5, both polarizations are likely to be delayed and one can try to change this delay by using an optical delay, electrical delay or buffer on the path of one of the data transmitted on one of the multiplexed polarizations. Using the FEC information is possible in order to correct the delay between polarizations. In case of optical polarization demultiplexing, this can be done using an optical variable delay line on the path of the light signal of the polarization to be controlled, or an electrical phase shifter or a buffer on the path of the electrical signal of the polarization to be controlled. 
     In case of polarization demultiplexing through digital signal processing, it is possible to place a buffer or a filter to delay the data of the polarization to be controlled, as explained in “”Digital Communication Receivers” (Wiley-Interscience Publication, 1998, p 505) by H. Meyr et al. However, in all cases, the information given by the result on the FEC frame monitoring is binary in the sense that if a delay appears, there is no information on the delay amount or on which polarization is delayed. 
     Therefore, correcting the delay between polarization-multiplexed signals with this method is likely to require several attempts before finding the adequate delay compensation, complicating the practical implementation. Moreover, continuous control of the delay between polarized signals is not implementable with this method. Finally, using this method requires to use the information of upper layer of the network to control physical layer parameters, which may complicate system design or hinder interoperability. 
     Another approach is particular to digital polarization demultiplexing. In this case, a coherent receiver is used and the resulting signals are converted from analog to digital. The converted signals are processed in order to recover the transmitted data. During the signal processing, the signals for each polarization are recovered through algorithms such as Constant Module Algorithm (C.M.A.). This is explained by N. Kaneda et al. in “Coherent Polarization-Division-Multiplexed QPSK Receiver with Fractionally Spaced CMA for PMD Compensation” (Photonics Technology Letter, Vol. 21, No. 4, Feb. 15, 2009). 
     Digital processing enables to process the signal through digital filters, which enables to correct some of the degradations of the signal appearing during the transmission through the medium. Filters can be updated through the recognition of training sequences used at regular interval during the transmission, as described in “Ultra Long-Haul QPSK Transmission using a Digital Coherent Receiver” (LEOS  2007 ) by S. J. Savory et al. Training patterns can also be used to eliminate the ambiguity on polarizations and transmitted symbols. All possibilities for the attribution of ambiguous variable are tried, until the training pattern is recognized. 
     In this case, buffer or time delaying filters can be used to correct the delay between polarizations until the training pattern is recognized, or the received pattern can be compared with several recognition patterns, for possible cases of delay, until there is a match, which enables to retrieve and correct the delay between polarizations. This approach requires training patterns, which are susceptible to decrease the amount of transmitted data at constant bit rate. Moreover, this method requires one comparison for each case of delay. Therefore it multiplies the calculation time and required memory by the number of cases to study. This will consummate more electrical power for the increased computation requirements. 
     However, there is room for improvement in simplicity, possibility of continuous control, processing power consumption, monitoring range and speed for monitoring the delay between polarization multiplexed signals. 
     Parallel transmission can be implemented with WDM. The parallel lower rate channels are transmitted by lightwave carrier at different wavelengths through the same medium. Each channel is emitted by a transmitter and the light of all transmitters are multiplexed into the same fiber. At the other end of the link, the wavelengths are demultiplexed according to their wavelengths; each channel is received and decoded by receivers. The electrical data from the parallel receivers are then converted from parallel to serial. Intra channel skew can appear inside the medium due to chromatic dispersion or after reception due to difference of length of the electrical path. Dynamic network management and switching can change dynamically the intra-channel skew of received signals as the transmission path is changed dynamically. With higher bit rate, the symbol period decreases, therefore the skew problem becomes critical. This is illustrated in “Terabit LAN with optical virtual concatenation for Grid applications with super-computers” by M. Tomizawa et al. (OFC 2005 OThG6). 
     The presented solution rectify the skew between channels is the use of the XAUI standard, which relies on channel decoding and realigning. This requires an increase of the total bit rate for the same transmitted data payload, as the prefix needed for alignment is introduced in the transmitted data. In addition, the skew monitoring requires the information of upper layer information. 
     There is a room for improvement in term of simplicity, efficiency of the transmitted data rate. 
     Another way of implementing parallel transmission is SDM. The use of fiber ribbon for SDM and the skew problem inherent to SDM is illustrated in “All Optical Bit Parallel Transmission Systems” by A. P. Togneri et al. (SMBO IEEE 2003). The link skew is due to difference in the fiber length or condition. With higher bit rates, the symbol rate transmitted through the fiber increases, meaning shorter symbol periods. Therefore the skew problem becomes critical with higher bit rates. US Patent US2000484961A discloses a method to compensate skew by decoding and realigning data. This method requires the use of prefixes, which necessitates an increase of the total bit rate for the same transmitted data payload. Moreover, this method requires the use of information from higher layers than the physical layer. 
     There is room for improvement in term of simplicity, efficiency of the transmitted data rate. There is a need for a fast, simple method to monitor skew between polarization parallel channels on wide skew ranges, without relying on the information of upper layers, without the presence of a training sequence or prefix. 
     In conjunction with the above description, Japanese Patent Application Publications (JP-P2003-218844A, JP-P2004-193817A, and JP-A-Heisei 11-341102) enable to monitor and correct the skew between parallel channels. 
     Japanese Patent Application Publication (JP-P2003-218844A) requires that a special pattern named PING or PONG depending on the case is added to the actual data to be transmitted. PING and PONG pattern carry no information and are used only for the purpose of skew monitoring or compensation. Therefore, to carry effectively m bits of data within the network, JP-P2003-218844A requires in fact m bits plus the number of bits contained in PING or PONG to be physically transmitted through the network. Therefore, the bandwidth of the transmitters and receivers in the network when JP-P2003-218844A is implemented has to be higher than the bandwidth necessary to carry the data. This increases the cost and complexity of receivers and can cause a degradation of the performance of the receiver. Moreover, in JP-P2003-218844A, the receiver must distinguish PING and PONG patterns from the data. Therefore, PING and PONG patterns must be composed so they cannot be mistaken for data and data must be encoded so that it cannot be mistaken for PING or PONG pattern. In addition, measuring skew between channels in JP-P2003-218844A requires that the data received is first decoded so that PING and PONG patterns can be read and compared. When errors appear due to noise or degradation of the signal transmitted through the network, this can affect read PING and PONG patterns, and therefore cause an error in the skew evaluation. As a consequence, this can have a tremendous impact on the serialized signal that FEC or other correction method may not be able to compensate. Finally, in JP-P2003-218844A, the skew is measured in shifts of bits or symbols of PING and PONG patterns transmitted through the network. This is a coarse estimation which is limited to a one symbol period resolution. 
     Also, Japanese Patent Application Publication (JP-P2004-193817A) requires that special patterns named I(n), where n is an integer indexing the parallel channel where the pattern is inserted, are added to the actual data to be transmitted. I(n) patterns carry no information and are used only for the purpose of skew monitoring or compensation. Therefore, to carry effectively m bits of data within the network, JP-P2004-193817A requires in fact m bits plus the number of bits contained in I(n) patterns to be physically transmitted through the network. Therefore, the bandwidth of the transmitters and receivers in the network when 2004-193817 is implemented has to be higher than the bandwidth necessary to carry the data. This increases the cost and complexity of receivers and may cause a degradation of the performance of the receiver. Moreover, in JP-P2004-193817A, the receiver must distinguish I(n) patterns from the data. Therefore, I(n) patterns must be composed so they cannot be mistaken for data and data must be encoded so that it cannot be mistaken for I(n) patterns. In addition, measuring skew between channels in JP-P2004-193817A requires that the data received is first decoded so that I(n) patterns can be read and compared. In cases where errors appear due to noise or degradation of the signal transmitted through the network, this can affect the reading of I(n) patterns, and therefore cause an error in the skew evaluation. As a consequence, this can have a tremendous impact on the serialized signal that FEC or other correction method may not be able to compensate. Finally, in JP-P2004-193817A, the skew is measured in shifts of bits or symbols of I(n) patterns transmitted through the network. This is a coarse estimation which is limited to a one symbol period resolution. 
     Also, Japanese Patent Application Publication (JP-A-Heisei 11-341102) requires that m′ frame bits are added to the actual data to be transmitted. Therefore, to carry effectively m bits of data within the network, H11-341102 requires in fact m+m′ bits to be physically transmitted through the network. Therefore, the bandwidth of the transmitters and receivers in the network when JP-A-Heisei 11-341102 is implemented has to be higher than the bandwidth necessary to carry the data. This increases the cost and complexity of receivers and may cause a degradation of the performance of the receiver. Moreover, in JP-A-Heisei 11-341102, the receiver must distinguish m′ bits of the frame from the data. Therefore, the frame must be especially composed so it cannot be mistaken for data and data must be encoded so that it cannot be mistaken for the frame. Other proposed methods of generating the frame requires information from layers higher than the physical layer of the network in order to monitor and compensate skew to the channel. Such requirement increases the complexity of the receiver. Skew monitoring is dependant on higher layer information and in return, higher layer information necessitates skew compensation to be retrieved, and skew compensation depends on the monitored skew. This may have a negative impact on the stability of the integrity of the received data when distortions appear on the transmission path. Finally, in JP-A-Heisei 11-341102, the skew is measured in shifts of bits or symbols of the frame transmitted through the network. This is a coarse estimation which is limited to a one symbol period resolution. 
     SUMMARY OF INVENTION 
     The object of the present invention is to provide a simple method to monitor the transmission skew between parallel signals generated from a serial signal, usable on wide skew ranges and without necessitating the information of upper layers, or the presence of a training pattern or prefix added to the transmitted data. 
     In an aspect of the present invention, a method is provided with: monitoring the skew between parallel signals. The monitoring step includes: imprinting dips on the optical amplitude of the light signal between a determined number of symbols for each parallel signal; detecting electrical signals corresponding to the optical amplitude dips of the parallel channels to monitor; generating a signal proportional to the transmission skew from said electrical signals. Said dips are carved between symbols of the light signal and preferably do not affect the central part of the symbol, therefore do not affect the quality of the transmitted information. In another aspect of the invention, the period of the optical amplitude dip is different for each parallel signal and the frequency of the detected electrical signals are converted in order to be compared and to generate the information proportional to the intra-channel skew. 
     In another aspect of the invention, the period of the optical amplitude dip is different for each parallel signal and the frequencies of the detected electrical signal are converted to be compared. Moreover, the difference of the frequency of the dips is used to distinguish the parallel signals at the receiver. 
     In another aspect of the invention, the electrical signals containing the information of the dips are recovered for each parallel signal after the demultiplexing of the parallel signal. 
     In another aspect of the invention, the period of the optical amplitude dip imprinted on the parallel signals is changed during the monitoring of the transmission skew, in order to improve the precision of the monitoring. The generated signal proportional to the skew is generated for different cases of period of the optical amplitude dip. 
     In another aspect of the invention, the electrical signals recovered for the parallel signals are used to generate a clock signal. Said clock signals can be used to control the demultiplexing of the parallel signals or the processing of the received signals. 
     In still another aspect of the invention, an optical transponder is provided with an optical transmitter emitting polarization-multiplexed signals carrying information to transmit and an optical receiver recovering the data carried by polarization-multiplexed signals. The transmitter is provided with an apparatus capable of imprinting dips on the optical amplitude between a fixed number of symbols for each polarization multiplexed signal. The receiver is provided with: a polarization demultiplexing scheme; apparatus capable of detecting an electrical signal related to the optical amplitude dip for each polarization-multiplexed signal; apparatus generating a signal proportional to the transmission skew between the polarization multiplexed signals from said electrical signal. 
     In still another aspect of the invention, an optical transponder is provided with: optical transmitters emitting parallel signals at different wavelengths generated from a higher bit rate serial signal; optical receivers recovering the data carried by the parallel signals. The transmitters are provided with apparatus capable of imprinting dips on the optical amplitude between a fixed number of symbols for each parallel signal. The receivers are provided with: apparatus capable of generating electrical signals related to the optical amplitude dip for each parallel signal; apparatus capable of generating a signal proportional to the transmission skew between the wavelength multiplexed signals from said electrical signals. 
     In still another aspect of the invention, an optical transponder is provided with: optical transmitters emitting parallel signals generated from a higher bit rate serial signal, where the parallel signals are space division multiplexed; optical receivers recovering the data carried by the parallel signals. The transmitters are provided with apparatus capable of imprinting dips on the optical amplitude between a fixed number of symbols for each parallel signal. The receivers are provided with: apparatus capable of generating electrical signals related to the optical amplitude dip for each parallel signal; apparatus generating a signal proportional to the transmission skew between the space division multiplexed signals from said electrical signals. In still another aspect of the invention, a measurement system is provided with an apparatus capable of imprinting an amplitude variation between a fixed number of symbols for each parallel signal, and another apparatus capable of detecting an electrical clock signal related to said optical amplitude variation for each polarization-multiplexed signal, detecting the phase difference between said clock signals for the polarization-multiplexed signals. The system is capable of returning a value proportional to the phase difference between the clock signals. 
     In the present invention, dips are inserted between a fixed number of symbols, but the time period of the symbols or the number of symbols to be transmitted within the network does not change. Therefore, when implementing the present invention, the transmitters and receivers can have the same bandwidth as the one necessary to carry the actual data, which is an improvement when compared to background art relying on overheard or training patterns. In addition, in the present invention, no constraint is added to the encoding of the data, which makes encoding more simple when compared to the use of overheads or training patterns. In the present invention, the skew monitoring does not required data to be decoded as with overheads or training patterns, therefore it is not directly sensible to errors within transmitted bits. The present invention compares electrical signals generated from the detected dips, therefore the measurable skew is finer than one bit or symbol period transmitted through the network. In the present invention, no constraint is added to the relation between the physical receiver and higher layer information. Skew monitoring is independent from higher layer information, which enable simpler receiver structures as compared to background art using FEC or upper layer information. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
       The above and other advantages and features of the present invention will be more apparent from the following descriptions taken in conjunction with the accompanied drawing, in which: 
         FIG. 1  is a block diagram schematically showing a configuration of an apparatus generating optical amplitude dips; 
         FIG. 2  is a diagram showing time charts of the apparatus of  FIG. 1 ; 
         FIGS. 3A to 3D  are diagrams showing simulation results for a light signal with dips imprinted on the optical amplitude; 
         FIG. 4  is a block diagram schematically showing a configuration of an apparatus generating a signal proportional to a delay between clock signals; 
         FIGS. 5A to 5E  are diagrams showing time charts and a graph of the apparatus of  FIG. 4 ; 
         FIG. 6  is a block diagram schematically showing a configuration of an optical transmitter emitting polarization multiplexed signals with dips imprinted on the light signals; 
         FIG. 7  is another block diagram schematically showing a configuration of the optical transmitter emitting polarization multiplexed signals with dips imprinted on the light signals; 
         FIG. 8  is a block diagram schematically showing a configuration of a transponder emitting and receiving polarization multiplexed signals in case of application to digital demultiplexing of the received signals; 
         FIG. 9  is a schematic representation of a transponder emitting and receiving polarization multiplexed signals, implementing the invention applied to digital demultiplexing of the received signals; 
         FIG. 10  is a schematic representation of a transponder emitting and receiving polarization multiplexed signals, implementing the invention applied to optical demultiplexing of the received signals; 
         FIG. 11  is a schematic representation of a transponder emitting and receiving wavelength division multiplexed signals and implementing the invention; 
         FIG. 12  is a schematic representation of a transponder emitting and receiving space division multiplexed signals and implementing the invention; and 
         FIG. 13  is a schematic representation of a measurement instrument implementing the invention. 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
       FIG. 1  is a block diagram schematically showing a configuration of an apparatus, which can be used in order to imprint a dip on the optical amplitude of a light signal. Before explaining in detail the apparatus of  FIG. 1 , the Pseudo-Return-to-Zero format of index n (P-RZ(n)) is defined in analogy to Return-to-Zero (RZ) and Non-Return-to-Zero (NRZ) formats. In RZ format, the optical amplitude returns to zero between two consecutive symbols. In NRZ format, the optical amplitude does not return to zero between consecutive symbols, and it is forced to zero if and only if the symbol amplitude is zero or crosses zero. In P-RZ(n) format, the amplitude returns to zero between consecutive symbols every n symbols and does not return to zero in the other cases, unless the symbol amplitude is zero or crosses zero. P-RZ(n) format is not a RZ format of at an n time slower clock frequency, as the amplitude is forced to zero and is relaxed at a very steep slope (same slope as RZ). P-RZ(n) causes dips on the optical amplitude every n symbols, as required by the present invention and is therefore suitable for the present invention. On the contrary, RZ format at an n time slower clock frequency would cause a n time slower slope when the amplitude is forced to zero and would cause degradation of the quality of the transmission as the central part of symbols are affected by the slow slope of the amplitude change. Therefore, RZ at an n time slower clock speed is not suitable for the present invention, whereas P-RZ(n) is a valid option. As an extension of our notation, P-RZ( 1 ) is RZ format and P-RZ(∞) is NRZ format. 
     A system  100  is a P-RZ(n) carver, which transforms a NRZ light signal  101  into a P-RZ(n) light signal  102 . A clock signal  103  is a signal at the symbol rate (of frequency f) of the input NRZ light signal  101 . The clock signal  103  is divided by a divider  120  into two identical clock signals. One of these signals is divided again by a divider  121 , to be fed into D-type Flip Flops (DFF)  140  and  141 . The other output signal of the divider  120  is transformed by a frequency divider  130  into a clock signal  110  of frequency f/n. The DFF  140  detects the signal  110  at a rising edge according to the clock signal of frequency f supplied from the divider  121 . The output of flip-flop  140  is a signal  111 , which is divided by a divider  122 . One of the outputs of the divider  122  is fed into a logical NAND gate  142 , and the other output of the divider  122  is fed into the DFF  141 . The DFF  141  detects the output of the divider  122  (equal to the signal  111 ) according to the clock signal of frequency f supplied from the divider  121 . The clock input of the D-type flip-flop  141  is negated, so that the D-type flip-flop  141  operates on the falling edge of the output of the divider  121 . The negated output of the D-type flip-flop  141  is a signal  112 , which is fed into the logical NAND gate  142 . The output signal of the gate  142  is a signal  113 , which is fed into a driver  151 , which converts it into a voltage to command an intensity modulator  150 , so that the transmission of the intensity modulator  150  becomes maximum when the signal  113  is high and the transmission of the intensity modulator  150  becomes minimum (ideally null) when the signal  113  is low. 
       FIG. 2  is a diagram showing timing charts of logical signals used in the P-RZ(n) carver shown in  FIG. 1 , when n=4. In  FIG. 2 ,  200  denotes the clock signal  103  of the frequency f, and  201  denotes the clock signal  110  of the frequency f/4.  202  denotes the output  111  of the D-type flip flop  140 ,  203  denotes the output  112  of the D-type flip flop  141 , and  204  denotes the output  113  of the NAND gate  142 . In this case, dips of f/2 width are imprinted on the light signal  101  every other 4 symbol. In that case, the output  102  is P-RZ( 4 ) signal. According to  FIG. 2 , the apparatus of  FIG. 1  enables to generate P-RZ(n) format, which is suitable for the present invention. 
       FIGS. 3A to 3D  represent graphs of simulated results for different QPSK transmitters. All the transmitters have the bit rate of 56 GHz, i.e. the baud rate of 28 Gbaud (f=28 GHz according to the notation of the explanation of  FIG. 1 ). A graph  300  represents an optical amplitude for a NRZ QPSK (P-RZ(∞)-QPSK) transmitter and the corresponding constellation map  310  is plotted. The optical amplitude is a maximum (1.4 AU) during at the center of symbols and between symbols when consecutive symbols have the same phase. The optical amplitude drops to 1 AU between symbols when consecutive symbols have a phase difference of (±π/2) radians. The optical amplitude is minimum (0 AU) between symbols when consecutive symbols have a phase difference of π radians. A graph  301  represents the optical amplitude for a RZ-QPSK (P-RZ( 1 )-QPSK) transmitter and the corresponding constellation map  311  is plotted. The amplitude drops to 0 AU between consecutive symbols, whatever the phase difference between symbols may be. A graph  304  represents the optical amplitude for a P-RZ( 4 )-QPSK transmitter and the corresponding constellation map  314  is plotted. The optical amplitude is similar to one NRZ-QPSK, but with a dip imprinted every 4 symbols. The dip imprinted by P-RZ( 4 ) is identical to the dip of RZ between symbols. A graph  308  represents the optical amplitude for a P-RZ( 8 ) QPSK transmitter and the corresponding constellation map  318  is plotted. The optical amplitude is similar to one NRZ QPSK, but with a dip imprinted every 8 symbols. The dip imprinted by P-RZ( 8 ) is identical to the dip of RZ between symbols. A spectrum  320  of the electrical signal generated through the conversion of the light signal of the NRZ-QPSK transmitter by a photo-diode. One peak is standing at the symbol frequency (f=28 GHz). A spectrum  321  of the electrical signal generated through the conversion of the light signal of the RZ QPSK transmitter by a photo-diode. One peak is standing at the frequency f. A spectrum  324  of the electrical signal generated through the conversion of the light signal of the P-RZ( 4 ) QPSK transmitter by a photo-diode. The main peak stands at the frequency f/4, followed by another peak at f. Lesser peaks are presents at other harmonics of the frequency f/4. A spectrum  328  of the electrical signal generated through the conversion of the light signal of the P-RZ( 8 ) QPSK transmitter by a photo-diode. The main peak stands at the frequency f, followed by another peak at f/4. Lesser peaks are presents at other harmonics of the frequency f/8.  FIG. 2  illustrates the fact that for P-RZ(n), a dip is carved every n symbols. When converting the light signal into electrical signal, the main frequency components appear at f and f/n. 
       FIG. 4  is an apparatus which can be used to measure a phase difference between two signals at the same frequency. A phase comparator  400  compares a phase difference between input signals  401  and  402  and generates a voltage  403 , which is proportional to the phase difference. Two toggled flip flops  424  and  425  have clear terminals. The input signals  401  and  402  are connected to the clock inputs of the flip flops  424  and  425 , respectively. J inputs of the flip flops  424  and  425  and inverted PR bar inputs of the flip flops  424  and  425  are connected to a high level voltage  410 . Inverted K bar inputs of the flip flops  424  and  425  are connected to the ground levels  411  and  412 . Q outputs  404  and  405  of the flip flops  424  and  425  are connected to the inputs of a logical NAND gate  420 . The output  406  of the NAND gate  420  is connected to the inverted CLR bar inputs of the flip flops  424  and  425 . Integrating amplifiers  431  and  432  amplify and integrate the signals  404  and  405  over one period of the input signals  401  and  402 , respectively 
       FIGS. 5A to 5D  represent time charts of signals  401 ,  402 ,  404 ,  405  and  406  for different phase difference cases, and  FIG. 5E  is a graph. In  FIG. 5A ,  501 ,  502 ,  504 ,  505  and  506  represent time charts of the signals  401 ,  402 ,  404 ,  405  and  406 , respectively, when the phase difference between the clock signals  401  and  402  is 0, and when the first clock falling edge transition is detected on the clock signal  401 . In  FIG. 5B ,  511 ,  512 ,  514 ,  515  and  516  represent time charts of the signals  401 ,  402 ,  404 ,  405  and  406  respectively, when the phase difference between the clock signals  401  and  402  is π/8, and when the first clock falling edge transition is detected on the clock signal  401 . In  FIG. 5C ,  521 ,  522 ,  524 ,  525  and  526  represent time charts of the signals  401 ,  402 ,  404 ,  405  and  406  respectively when the phase difference between the clock signals  401  and  402  is π, and when the first clock falling edge transition is detected on the clock signal  401 . In  FIG. 5D ,  531 ,  532 ,  534 ,  535  and  536  represent time charts of the signals  401 ,  402 ,  404 ,  405  and  406  respectively when the phase difference between  401  and  402  is 7π/8, and when the first clock falling edge transition is detected on the clock signal  401 . As shown in  FIG. 5E , a graph  590  plots an output signal  407  of the integrating amplifier  431  against the phase difference between the clock signals  401  and  402 , when the first clock falling edge is detected on the signal  401 . In that case, an output signal  408  of the integrating amplifier  432  is null. When the first clock falling edge is detected on the signal  402 , the graph  590  can be seen as a plot of the signal  408  plotted against the phase difference between the signals  402  and  401 , the signal  407  being null. This is due to the symmetry between elements  401 ,  424 ,  404 ,  420  on one side and elements  402 ,  425 ,  405 ,  420  on the other side. A processing circuit  430  detects the signals.  407  and  408 . If the signal  408  is lower than the signal  407 , this means that the first clock falling edge was detected on the signal  401  and the processing circuit  430  generates a signal  403  which is equal to the signal  407 . If the signal  407  is lower than the signal  408 , this means that the first clock falling edge was detected on the signal  402  and the processing circuit  430  generates the signal  403  which is a difference between a maximum value achievable by the signal  408  and a current value of the signal  408 . Therefore, the signal  403  is proportional to the phase difference between the signals  401  and  402  in all cases. The apparatus  400  can generate the voltage  403  proportional to a delay between the signals  401  and  402 . 
       FIG. 6  is a block diagram schematically showing an optical Polarization Multiplexed (PM) QPSK transmitter emitting polarization multiplexed signals with dips of different frequencies imprinted on each polarization signal. An optical Polarization Multiplexed P-RZ(l,m) QPSK transmitter  600  emits two PM light signals, one being P-RZ(l)-QPSK and the other P-RZ(m)-QPSK. A serial electrical data  601  is coded and de-serialized into two parallel electrical signals  603  and  604 . The parallel signals  603  and  604  are converted into appropriate voltages by drivers  611  and  612  to imprint a data on the phases of optical carrier signals  605  and  606  by QPSK modulators  631  and  632 . A clock signal  609  of a frequency f equal to a baud rate of an emitted optical carrier signal  602  is split and divided by l and by m by frequency dividers  613  and  614 . The clock signals at frequency f/l and f/m are fed into P-RZ(l) and P-RZ(m) carvers  633  and  634 , which are identical to the carver  100  of  FIG. 1 . A laser  620  emits an optical carrier signal linearly polarized, which is divided by a polarization maintaining coupler  621  with the ratio of 50:50 into the two optical carrier signals  605  and  606 . The P-RZ(l) carver  633  imprints dips at the frequency f/l on the optical carrier signal  605 . The P-RZ(m) carver  634  imprints dips at the frequency f/m on the optical carrier signal  606 . The phases of the optical carrier signals  605  and  606  are modulated by QPSK modulators  631  and  632 . A polarization rotator  636  rotates the polarization of the carrier signal  606  by π/2 radians, so that the polarizations of modulated light signals  607  and  608  are orthogonal to each other. The orthogonal polarizations  607  and  608  are multiplexed by a polarization combiner  622 . The light signal  602  is a result of the polarization multiplexing of two parallel signals carrying the de-serialized information of the serial signal  601 . One polarization has dips on the optical amplitude at the frequency f/l. The other polarization has dips on the optical amplitude at the frequency f/m. 
       FIG. 7  is a block diagram schematically showing an optical Polarization Multiplexed QPSK transmitter emitting polarization multiplexed signals with dips imprinted on each polarization signal. A serial electrical data  701  is fed into an optical Polarization Multiplexed P-RZ(k) QPSK transmitter  700 . A light signal  702  is emitted by the transmitter  700 . Signals  703 ,  704 ,  705 ,  706 ,  707 ,  708  and  709  are similar to the signals  603 ,  604 ,  605 ,  606 ,  607 ,  608  and  609  in  FIG. 6 . Devices  710 ,  711 ,  712 ,  720 ,  721 ,  722 ,  731 ,  732  and  736  are identical to the devices  610 ,  611 ,  612 ,  620 ,  621 ,  622 ,  631 ,  632  and  636  in  FIG. 6 . A variable optical delay line  735  has been placed on the optical path between the modulator  731  and the polarization combiner  722  in order to shift the polarization before the emission. The clock signal  709  of the frequency f equal to the baud rate of the emitted lightwave signal  702 , passes through a frequency divider  713  and the resultant clock signal of the frequency f/k is fed into a P-RZ(k)  723 , which is similar to the carver  100  on  FIG. 1 . The carver  723  imprints dips on the optical amplitude of the signal emitted by the laser  720  and the resultant carved signal is split by a polarization maintaining coupler  721 . The light signal  702  is the result of the polarization multiplexing of two parallel signals carrying the de-serialized information of the serial signal  701 . Both the polarizations have dips on the optical amplitude of the frequency f/k. The variable delay line  735  enables to shift the timing of the emitted multiplexed signals. For example, in one configuration, the dips for each polarization coincide (bit superposed). In another configuration, the dips of one polarization coincide with the middle of an interval between dips of the other polarization, which is the delay of k/2 symbols (bit interleaved). 
     In one exemplary embodiment, the present invention is incorporated in a transponder shown in  FIG. 8 . A polarization multiplexed QPSK transponder  800  incorporates the present invention. The transponder  800  has an optical transmitter  801 , which is similar to the transmitter  600  in  FIG. 6 , and an optical receiver  802 . A 100 Gbit/s data stream  810  is de-serialized and transmitted by the transmitter  801  on two polarizations of a light carrier signal  811 , at the symbol rate of 25 Gbaud, therefore, at the symbol period of 40 ps. We select l=8 and m=4 in order to enable a path skew monitoring of ±40×(m/2)=80 ps. 
     Another polarization multiplexed QPSK signal  812  at 25 Gbaud emitted by an optical transmitter similar to the transmitter  801 . The receiver  802  receives the signal  812 , monitors a transmission skew of the signal  812  and compensates it, decodes and re-serializes the tributary signals into the 100 Gbit/s data stream  813 . The operation principle of the transmitter  801  has already been described with reference to  FIG. 6 . 
     Now, an example of operation of  802  will be described. A coherent receiver  821  consists of two 90° hybrids mounted in a polarization diversity scheme, their output being connected to four differential photodiodes (not shown, which are included in the receiver  821 ), each of them being connected to an amplifier (which is also included in the receiver  821 ). The receiver  821  receives the signal  812 , mixes it with the light signal from a laser  820  used as a local oscillator, and converts lightwave signal into four electrical signals. The output signals of the receiver  821  are converted into four digital signals by the Analog to Digital Converter (ADC) Unit  822  operating at 2 samples by symbol per signal. The resultant four digital signals are transmitted to a processing unit  830 . 
     The polarizations, Polarization X and Polarization Y, are then demultiplexed in a demultiplexing unit  832 , where the processing is based on Constant Modulus Algorithm (C.M.A). A complex signal of one of the polarizations, the Polarization X, is then processed by a unit  840  to calculate its squared modulus, which reflects the properties of the optical amplitude of the received signal. The spectrum of the square modulus is calculated by Fast Fourier Transform (FFT) by an FFT unit  842  and the result is analyzed by an analyzing unit  844 , which looks for a component of f/1 (3.125 GHz) in order to check if the Polarization X is a signal having the P-RZ( 8 ) component and not the signal having the P-RZ( 4 ) component. If the signals demultiplexed at the demultiplexing unit  832  are inverted, the unit  844  does not find the f/1 component and the unit  844  instructs the unit  832  to invert the outputs of both of retrieved polarization signals, Polarization X and Polarization Y. The use of the component of P-RZ( 8 ) to identify the signals enables to reduce the identification cases of a prefix or training pattern incorporated in the transmitted light signal. In a similar manner, the square modulus of Polarization Y is calculated by a unit  841 , its spectrum is analyzed by a FFT unit  843  and the component of f/m=6.25 GHz can be used to retime the data in a clock generator  823 . The output of a unit  833  is then used by a clock generator  823  to generate the clock signal for the timing of the ADC unit  822 . 
     The component of P-RZ( 4 ) is used for clock recovery in the processing unit  830 , with modest processing resource. A delay monitor unit  848  analyzes the square modulus of each of Polarization X and Polarization Y and calculates a delay between dips of f/m=2×f/l=6.25 GHz. The delay monitor unit  848  calculates a link skew between the multiplexed polarizations from the P-RZ carvers of the transmitter, which emits the signal  813 , to the coherent receiver  820 . The unit  848  detects the skew between the polarizations X and Y by comparing the information from the units  840  and  841 . A delay control processing unit  849  controls delay filtering units  846  and  847  according to the information from the delay monitor unit  848  to filter Polarization X and Polarization Y with buffers and interpolation filters in order to compensate the skew between Polarization X and Polarization Y. That is, the delay control processing unit  849  computes the coefficients and buffer size of the units  846  and  847 , according to the skew detected by the unit  848  in order to compensate for the skew between the polarizations X and Y. Units  834  and  835  are identical and may be integrated. CPE stands for Carrier Phase Estimation. It is described in detail in “PLL-Free Synchronous QPSK Polarization Multiplex/Diversity Receiver Concept With Digital I&amp;Q Baseband Processing” (IEEE Photonics Technology Letters, VOL. 17, NO. 4, April 2005 pp 887-889) by R. Noe. In short, it is a calculation method, which enables to recover phase information after coherent reception without using a PLL. Dec stands for Decoding. Therefore the role of the units  834  and  835  is to recover the phase information and to decode this phase information (generate binary data for the recovered received information) for the polarizations X and Y. The units  834  and  835  compensate for a deviation of the frequency of the optical signal from the laser  820  based on the frequency of the optical carrier signal of the signal  812  by a carrier phase estimation method, and decide the symbols detected for Polarization X and Polarization Y. A unit  836  serializes and decodes the signals of the multiplexed polarizations X and Y. The signal  813  is the 100 Gbit/s signal received by the receiver  802 . 
     The link skew between polarizations X and Y has been compensated by the receiver using the monitoring information of the present invention, so that the data signal  812  has been correctly received and converted into the data signal  813 . In addition, the present invention can perform the clock recovery of the received signal and the distinction between the polarization X and the polarization Y, with moderate processing requirement and power consumption. The monitoring of skew does not necessitate the information of upper layers. The monitoring of skew does not necessitate the implementation of a training sequence. 
     In another exemplary embodiment, the frequency dividers  613  and  614  can be configured to change the integers l and m, by which the clock signal of the P-RZ carver is divided. During the transmission, in a first phase, l and m can be set to be large integers, so that the monitoring range is very wide; in a second phase, l and m can be reduced to monitor on narrower ranges. In that case, the processing units  844  and  848  are able to adapt their operation to the change of l and m. 
     In another exemplary embodiment, the present invention is incorporated in the transponder described on  FIG. 9 . A polarization multiplexed QPSK transponder  900  incorporates the present invention. The transponder  900  has an optical transmitter  901 , which is identical to the transmitter  800  in  FIG. 8 , and an optical receiver  902 . The signals  910 ,  911 ,  912  and  913  are identical to the signals  810 ,  811 ,  812  and  813  in  FIG. 8 . The devices  920 ,  921 ,  922 ,  923  are identical to the devices  920 ,  921 ,  922  and  923  in  FIG. 8 . The processing units  932 ,  935 ,  936 ,  946 ,  947  and  949  are identical to the processing units  832 ,  835 ,  836 ,  846 ,  847  and  849  in  FIG. 8 . 
     In this embodiment, the correction of the link skew between the polarizations is performed analogically outside a processing unit  930 . First, the received light signal  912  is divided by a coupler  924 . One output of the coupler  924  is connected to the coherent receiver  921 . The other output is connected to a photodiode  950 , which transforms the optical amplitude of the signal  912  into an electrical signal. This signal is split in two and the split signals are filtered by a band pass filter  951  centered around f/m (6.25 GHz in the previous numerical examples) and a band pass filter  952  centered around f/l (3.125 GHz in the numerical example). The filtered signals are then two clock signals of the respective frequencies of 6.25 GHz and 3.125 GHz, which contain the P-RZ information of the respective Polarizations X and Y. The clock signals are respectively multiplied by a ratio of the least common multiple of both P-RZ indices and the opposite P-RZ index. In this case, the frequency multiplier  954  multiplies the clock signal after the filter  952  by 2 (=8/4) and the frequency multiplier  953  would multiply the clock signal after the filter  951  by 1. Thus, in this example, the filter  951  can be replaced by an electrical delay line. A phase comparator  955  is identical to the apparatus  400  in  FIG. 4 . The electrical paths for the clock signals are designed so that the output signal of the comparator  955  is proportional to the link slew between the polarizations X and Y of the signal  912 . The output signal of the comparator  955  is converted to a digital signal by an ADC circuit  956 . The information of the ADC circuit  956  is used to control the Delay Control units  946  and  947  inside the processing unit  930 . In addition, the clock signal of the frequency f/m after the filter  951 , can be used to control a clock generator  923  used to control the timing of an ADC unit  922 . It can also be used inside the processing unit  930 . The distinction of the X polarization and Y polarization is made by pattern recognition in the processing unit  930 . 
     The link skew between the polarizations X and Y has been compensated by the receiver using the monitoring information of the present invention, so that the data signal  912  has been correctly received and converted into a data signal  913 . The monitoring of the link skew does not necessitate the information of any upper layer. In addition, the present invention enabled the clock recovery of the received signal. 
     In another exemplary embodiment, the present invention is incorporated in the transponder described on  FIG. 10 . A polarization multiplexed DQPSK transponder  1000  incorporates the present invention. The transponder  1000  has an optical transmitter  1001 , which is identical to the transmitter  700  in  FIG. 7 , and an optical receiver  1002 . The modulation format is DQPSK, therefore a coder  710  codes the data signal according to the DQPSK format. The variable optical delay line  735  is set so that the polarization X has the delay of k/2 symbols from the polarization Y at the output of the transmitter. This is similar to a bit interleaved configuration for a RZ polarization multiplexed transmitter, and enables optical polarization demultiplexing. A signal  1010  is a 100 Gbit/s data stream, which is de-serialized and transmitted by a multiplexing unit  1001  on two polarizations of the light carrier signal  1011 , at the symbol rate of 25 Gbaud, therefore at the symbol period of 40 ps. We select k=4 in order to enable a path skew monitoring of ±40×(k/2)=80 ps. 
     A signal  1012  is another polarization multiplexed QPSK signal at 25 Gbaud emitted by an optical transmitter similar to the transmitter  1001 . The receiver  1002  receives the signal  1012 , monitors the transmission skew of the signal  1012  and compensates it, decodes and re-serializes the tributary signals into the 100 Gbit/s data stream  1013 . A polarization controller  1030  rotates the polarization of the signal  1012  so that a polarization beam splitter  1031  separates the multiplexed polarizations and sends each of them to couplers  1040  and  1041 . The polarizations are demultiplexed by one-symbol-delay interferometers  1032  and  1033  and the demodulated optical symbols are received by respective balanced photodiodes  1034  and  1035 . The balanced photodiodes  1034  and  1035  convert the light signal into electrical signals, and the data are recovered by Clock and Data Recovery (CDR) units  1036  and  1037 . The recovered data are then decoded and re-serialized by the parallel to a serial converter  1038 . The converter  1038  is able to identify and permute, if necessary, the polarizations. The coupler  1040  splits one part of the separated polarizations and sends it to the photodiode  1042 , which converts a signal proportional to the optical amplitude of the polarization into an electrical signal. The signal is filtered by a band pass filter  1044 , which extracts the information at the frequency f/4 of the separated polarization. The extracted clock signal is split. One part can be used for the feedback to the polarization controller  1030  through a control circuit  1039 , working as a maximum signal feedback. When the polarizations are perfectly demultiplexed, the clock signal is maximum as the dips carved by P-RZ( 4 ) on both polarizations do not coincide, due to the delay of k/2 symbols between them at the output of the transmitter. The other part of the clock signal after the filter  1044  is send into the phase comparator  1046 . The devices  1041 ,  1043  and  1045  are identical to the respective devices  1040 ,  1042  and  1044 . The clock signal of the frequency f/4 extracted by a band pass filter  1045  is sent to the phase comparator  1046 , which emits a signal proportional to the phase difference between the clock signals of the frequency f/4, therefore proportional to the skew between the polarizations. The phase comparator  1046  is identical to the apparatus  400  in  FIG. 4 . 
     A control circuit  1047  controls a differential polarization delay, which retards one polarization relatively to the other, so that the delay between the polarizations is maintained to k/2 symbols, as it is, when emitted by the transmitter, which emitted the signal  1012 . The retard of the X polarization by k/2 symbols is necessary for the optical process, and it is eventually corrected by a buffer of two symbols in the serializer and coder  1038  after the polarizations X and Y have been identified. The link skew between polarizations X and Y has been compensated by the receiver using the monitoring information of the present invention, so that the data  1012  has been correctly received and converted into the data  1013 . In addition, the present invention enabled to control the optical polarization demultiplexing scheme. 
     In another exemplary embodiment, the present invention is incorporated in the transponder described on  FIG. 11 . A WDM transponder  1100  incorporates the present invention. The transponder  1100  has an optical transmitter  1101 , which transmits a high bit rate data stream  1150  on four parallel signals at four different wavelengths and, whose bit rate are four time lower than the stream  1150 . The transponder  1100  has an optical receiver, which receives four parallel signals of four different wavelengths emitted by a transmitter similar to the transmitter  1101 , and serializes them into a serial data stream  1153 . A signal  1151  represents the four parallel WDM signals emitted by the transmitter  1101 . A signal  1152  represents the four parallel WDM signals received by a receiver  1102 . The bit rates of the stream signals  1150  and  1153  are equal. 
     A coder and de-serializer  1160  splits the stream signal  1150  into four parallel signals, which are sent to modulator integrated lasers  1110 ,  1120 ,  1130  and  1140 . Each of the lasers emits a wavelength NRZ intensity modulated signal at a distinct wavelength. A clock signal of the frequency f (one fourth of the bit rate of the de-serializer  1160 ) is frequency divided by p by a frequency divider  1161  (four instance p=4), and the clock signal of the frequency f/p is distributed to four P-RZ(p) carver similar to the carver  100  of  FIG. 1 . The P-RZ carvers are noted  1111 ,  1121 ,  1131  and  1141 . A wavelength multiplexer  1162  is used to combine the four P-RZ( 4 ) intensity modulated signals into the signal  1151 . 
     A wavelength de-multiplexer  1172  separates the four wavelengths of the signal  1152  into four monochromatic signals. The four signals are received by photodiodes  1112 ,  1122 ,  1132  and  1142 . The electrical signals generated by the photodiodes through the photoelectrical conversion of the parallel optical wavelengths of the signal  1152 . Each of the four electrical signals is split. Clock and data recovery (CDR) units  1113 ,  1123 ,  1133  and  1143  decode split parts of the electrical signals, and band pass filter  1114  centered on the frequency f/p filters the other split part of the electrical signal converted by the photodiode  1112 . Therefore, the clock signal of the frequency f/p at the output of the band pass filter contains the information of the P-RZ( 4 ) carving. The filters  1124 ,  1134  and  1144  are identical to the filter  1114  and operate in the same way to the signals from the respective photodiodes  1122 ,  1132  and  1142 . Phase comparators  1125 ,  1135  and  1145  are identical to the comparator  400  in  FIG. 4  and compare the clock signal  1114  and the respective clock signals  1124 ,  1134  and  1144  in phase. The electrical signals proportional to the phase delays between the clock signals are generated by the phase comparators  1124 ,  1134  and  1144  and sent to a phase shifter control unit  1171 . The phase shifter control unit  1171  generates four electrical signals to control the four variable phase shifters  1116 ,  1126 ,  1136  and  1146 , which are placed on the path of the data received by the respective photodiodes  1112 ,  1122 ,  1132  and  1142 . The control unit  1171  controls the phase shifters  1116 ,  1126 ,  1136 ,  1146  to compensate the skews between the four received wavelengths, which are monitored through the phase delay between the clock signals of the frequency f/p generated through the four band pass filters. The control unit  1171  is calibrated so that the four phase shifters introduce the same delay when the four received wavelength have no skew. The serializer and decoder  1170  transform the four parallel signals with compensated skew into the serial signal  1153 . 
     The link skew between the four wavelengths has been compensated by the receiver using the monitoring information of the present invention, so that the data signal  1152  has been correctly received and converted into the data stream  1153 . 
     In another exemplary embodiment, the present invention is incorporated in the transponder shown in  FIG. 12 . A SDM transponder  1200  incorporates the present invention. The transponder  1200  has an optical transmitter  1201 , which transmits a high bit rate data stream  1250  on ten parallel signals, whose bit rate are ten times lower than the signal  1250 , through ten different fibers which are attached in the fiber ribbon  1251 . The transponder  1200  has an optical receiver, which receives ten parallel signals of fibers attached with the fiber ribbon  1252  and emitted by a transmitter similar to the transmitter  1201 . The receiver  1202  serializes them into the serial data stream  1253 . The bit rates of the signals  1250  and  1253  are equal to each other. The bit rate of the parallel signals transmitted in the fiber ribbon  1251  is equal to the bit rate of the parallel signals transmitted in the fiber ribbon  1252 , and is one tenth of the bit rate of the signal  1250 . 
     The coder and de-serializer  1260  splits the signal  1250  into ten parallel signals. An emitter  1210  is composed of a laser, whose optical intensity is modulated with one of the parallel signals from the coder  1260 , and a P-RZ(q) carver  1221 . A frequency divider  1261  transforms a clock data of the frequency f (one tenth of the bit rate of the signal  1250 ) into a clock signal at the frequency f/q, which is fed to the P-RZ(q) carver  1221 . The emitter  1210  emits one light signal according to the parallel signal from the coder  1260  into the fiber ribbon  1251 . The nine other emitters noted by consecutive numbers from  1211  to  1219  are identical to the emitter  1210 , and transmits nine parallel intensity modulated P-RZ(q) signals containing the information of the nine other parallel signals from the coder  1260  into the nine remaining fibers of the ribbon  1251 . 
     A skew compensation receiver unit  1230  is composed of a photodiode  1222 , a CDR  1223 , a band pass filter  1224  centered on the frequency f/q and a variable phase shifter  1225 . The photodiode  1222  converts the light signal from one of the fibers of the signal  1252  into an electrical signal, which is split in two signals. One signal is send to the CDR  1223  to be converted to electrical data, and the other signal is filtered by the filter  1224  to generate a clock signal of the frequency f/q containing the harmonic of the frequency f/q of the P-RZ(q) light signal. The phase shifter  1225  shifts the data after the CDR  1223  and is control by a control unit  1271 . The other nine parallel signals transmitted by the fiber ribbon  1252  are received by the nine skew compensation receivers noted by nine consecutive numbers from skew compensation receiver units  1231  to  1239 , which are identical to the receiver unit  1230 . 
     The nine phase comparators noted from  1241  to  1249  compare in phase, the clock signal of the frequency f/q emitted by the receiver unit  1230  and the respective clock signals coming from the skew compensation receiver units  1231  to  1249 . The electrical signals proportional to the phase delay between the clock signals are generated by the nine phase comparators  1241  to  1249  and sent to the phase shifter control unit  1271 . The phase shifter control unit  1271  generates ten electrical signals to control the ten phase shifters  1225 , . . . of the receiver units  1230  to  1239 . The control unit  1271  controls the phase shifters to compensate the skew between the ten received signals, which are monitored through the phase delays between the clock signals of the frequency f/q generated by the band pass filters of the receiver units  1230  to  1239 . The control unit  1271  is calibrated so that the ten shifters introduce a same delay when the ten received parallel signals have no skew. The serializer and decoder  1270  transform the ten parallel signals with compensated skew into the serial signal  1253 . 
     The link skew between the ten parallel signals has been compensated by the receiver using the monitoring information of the present invention, so the data  1252  has been correctly received and converted into the data  1253 . 
     In another embodiment, the present invention is implemented in the measurement instrument  1300  illustrated in  FIG. 13. 1310 ,  1312  and  1315  denote optical input interfaces.  1311  and  1313  denote optical output interfaces. An electrical generator  1322  generates a clock signal of the frequency f. The P-RZ(k 1 ) carver  1320  imprints P-RZ dips to the signal input on the interface  1310  according to the clock signal from the generator  1322 . A frequency divider  1323  converts the clock signal of the frequency f into a clock signal of the frequency f/2 and imprints P-RZ dips with the carver  1321  on an light signal input at the interface  1312 . A delay between the interfaces  1310  and  1311  is designed and adjusted to be equal to the delay between the interfaces  1312  and  1313 . The photodiode  1330  converts the light signal input at the interface  1315  into an electrical signal split to be filtered by the two band pass filters  1332  and  1333 , centered respectively on the frequencies f and f/2. A frequency doubler  1335  converts the clock signal of the output of the filter  1333  into a clock signal of the same frequency as the one at the output of the filter  1332 . A phase comparator  1340  measures the delay between the clock signals from the filter  1332  and the frequency doubler  1335 . The result of the phase comparison is displayed and saved on a medium by an interface  1350 . 
     Next, an example of operation of  1300  will be described. The instrument  1300  is used to measure the slew between the polarizations in the transmitter  600 . The modulators  631  and  632 , as well as the carvers  633  and  634  are turned off, so that the output signal of the transmitter  600  is constant. The output fiber  602  is connected to the interface  1310 , and the interface  1311  is connected to the input fiber  601 . The output fiber  602  is connected to the interface  1312 , and the interface  1313  is connected to the input fiber  601 . The output fiber of the transmitter  600  is connected to  1315 . The carvers  1320  and  1321  imprint P-RZ dips on the two polarizations of the transmitter  600 . Two clock signals of the frequencies f and f/2 containing the information of the two imprinted P-RZ dips are generated through the carvers  1332  and  1333 . The clock frequency of the frequency f/2 is doubled by the frequency doubler  1335  and the two clock signals of the frequency f are compared by the phase comparator  1340 . The measured phase delay expresses the delay between the P-RZ dips, and therefore the skew between the polarizations. A user interface  1350  displays and saves to a medium the skew measured between the multiplexed polarizations in the transmitter  600 . By adjusting the fiber length or introducing a variable delay line, the skew at the output of the transmitter  600  can be set to zero after several measurements. The maximum measurable skew is 2/f. For instance, if f=1 GHz, the maximum measurable skew is 2 ns. 
     As described above, according to the present invention, various advantages can be obtained. Also, the above embodiments or examples may be combined a case of no contradiction. 
     While the present invention has been particularly shown and described with reference to the exemplary embodiments thereof, the present invention is not limited to these exemplary embodiments. It will be understood by those of ordinary skill in the art that various changes in form and details may be made therein without departing from the spirit and scope of the present invention as defined by the claims.