Abstract:
Differential signal detection circuitry with an integrated reference voltage. The reference voltage is added as an offset to the output voltage, and its integration ensures that variations in the reference voltage closely track variations in the signal. Accordingly, the detection threshold for the signal being detected remains more consistent over variations in the circuit manufacturing process, power supply voltage and operating temperature.

Description:
BACKGROUND 
       [0001]    1. Field of the Invention 
         [0002]    The present invention relates to differential signal circuits, and in particular, to circuits and methods for detecting differential signals with compensation for variations in circuit manufacturing processes, power supply voltage and operating temperature. 
         [0003]    2. Related Art 
         [0004]    Many systems, particularly differential signal systems, need to detect the amplitude of the input signal and compare it to the reference signal. By determining if the amplitude of the input signal is greater than or less than the predetermined reference, it can be determined whether an input signal is present. If no signal is present, remaining portions of the system can be idled or shut down to save power. Further, certain circuit operations need only be started upon detection of an input signal. 
         [0005]    As is well-known in the art, and discussed in more detail below, the basic stages of subject signal detection circuit are an input stage, a rectification stage (also discussed hereinbelow as a detection stage), a reference voltage generator stage and a comparator stage. The comparator compares the output of the rectification stage to the predetermined reference, e.g., a reference voltage. The input signal magnitude causing the comparator to change its output signal state is the trip point of the system and should, typically, equal the input signal magnitude sought to be detected. 
         [0006]    As is well-known, the output signal of the rectification stage and a reference voltage generator stage will vary over process (e.g., manufacturing process), power supply voltage and temperature (PVT). Additionally, the input stage often shows significant performance variations over PVT, and a significant contributor to these variations is the transconductance of the input stage devices. Accordingly, to correctly determine if the input signal is greater or less than the reference, the signal path (e.g., the input and rectification stages) and reference voltage generator stage need to track one another in order to minimize variations in the trip point as over PVT varies. 
         [0007]    One conventional technique to achieve such tracking has been done by sending the reference signals through a dummy signal path implemented as a replica of the input and rectification stages so that the input signal and reference signal see the same path. However, this consumes additional circuit area and power. 
         [0008]    Referring to  FIG. 1 , one example of a conventional rectification stage  10  can be implemented as shown. The differential input signal phases VINP, VINN are applied to the differential input amplifier formed by bipolar junction transistors Q 1 , Q 2 . Another transistor Q 3  is part of a common mode circuit implemented as a voltage follower circuit driven by a common mode voltage VCM (discussed in more detail below). Current sources  12 ,  14  provide equal biasing currents I. This results in load currents I 1 , I 2  being conducted by the load resistances RL, producing the positive OUTP and negative OUTN phases of the differential output signal. The rectification occurs due to the difference in voltages at the transistor emitters and the taking of the output at the collector electrodes. A difference current I 3  between the two circuit branches causes a voltage difference between the mutually connected emitters of the differential amplifier transistors Q 1 , Q 2  and the voltage follower transistor Q 3 . This voltage difference changes with the differential input voltage VID (=VINP−VINN), thereby causing the transistors Q 1 , Q 2 , Q 3  to operate with different emitter currents. This adds to the non-linear operation of the emitter follower circuit operation. Additionally, the output signal range is limited by the available range of output voltage across the load resistance RL. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0009]      FIG. 1  is a schematic diagram of a conventional differential signal detection circuit and its corresponding equations describing its operation. 
           [0010]      FIG. 2  is a schematic diagram of an improved differential signal detection circuit and its corresponding equations describing its operation. 
           [0011]      FIG. 3  depicts the circuit of  FIG. 2  and its corresponding equations describing its dependency on operating temperature. 
           [0012]      FIG. 4  depicts the circuit of  FIG. 2  along with its requirement of an external reference voltage stage for driving a comparator stage. 
           [0013]      FIG. 5  is a schematic diagram of differential signal detection circuitry with an integrated reference voltage in accordance with one embodiment of the presently claimed invention. 
           [0014]      FIG. 6  is a schematic diagram of differential signal detection circuitry with an integrated reference voltage in accordance with another embodiment of the presently claimed invention, along with an exemplary embodiment of an input stage. 
           [0015]      FIG. 7  is a schematic diagram of differential signal detection circuitry with an integrated reference voltage in accordance with another embodiment of the presently claimed invention and its corresponding equations describing its operation. 
           [0016]      FIG. 8  is a schematic diagram depicting the circuitry of  FIG. 7 , an input stage and a biasing voltage stage. 
           [0017]      FIG. 9  is a schematic diagram of the circuitry of  FIG. 9  plus circuitry for base current compensation. 
           [0018]      FIG. 10  is a functional block diagram of an exemplary embodiment of an integrated. circuit design and fabrication system operated in accordance with computer instructions. 
       
    
    
     DETAILED DESCRIPTION 
       [0019]    The following detailed description is of example embodiments of the presently claimed invention with references to the accompanying drawings. Such description is intended to be illustrative and not limiting with respect to the scope of the present invention. Such embodiments are described in sufficient detail to enable one of ordinary skill in the art to practice the subject invention, and it will be understood that other embodiments may be practiced with some variations without departing from the spirit or scope of the subject invention. 
         [0020]    Throughout the present disclosure, absent a clear indication to the contrary from the context, it will be understood that individual circuit elements as described may be singular or plural in number. For example, the terms “circuit” and “circuitry” may include either a single component or a plurality of components, which are either active and/or passive and are connected or otherwise coupled together (e.g., as one or more integrated circuit chips) to provide the described function. Additionally, the term “signal” may refer to one or more currents, one or more voltages, or a data signal. Within the drawings, like or related elements will have like or related alpha, numeric or alphanumeric designators. Further, to the extent that the figures illustrate diagrams of the functional blocks of various embodiments, the functional blocks are not necessarily indicative of the division between hardware circuitry. Thus, for example, one or more of the functional blocks may be implemented in a single piece of hardware. 
         [0021]    Referring to  FIG. 2 , an exemplary embodiment  100  of improved differential signal detection circuitry avoids the use of the load resistances RL and difference resistance R ( FIG. 1 ), and takes the output voltage signal OUTP-OUTN at the emitter electrodes. During transient signal operation, the base-emitter voltages VBE of the input transistors Q 10 , Q 11  and common mode transistor Q 12  will be equal for equal currents I from the current sources  102 .  104 . As indicated by the equations, the output voltage OUTP-OUTN is half of the input voltage VID/ 2 , and is linear over the range of input voltage. 
         [0022]    Referring to  FIG. 3 , notwithstanding its improved performance otherwise, this differential signal detection circuitry  100  has an offset voltage VBE 2 -VBE 1 , i.e., a difference between the base-emitter voltages of the input transistors Q 10 , Q 11  and the common mode transistor Q 12 , that is proportional to the transistor threshold voltage VT, which is dependent upon temperature. 
         [0023]    Referring to  FIG. 4 , and as discussed above, the differential output signal phases OUTP, OUTN must be compared to corresponding phases REFP, REFN of a differential reference voltage. An external reference generator  110  provides this reference voltage, which is used by a differential voltage comparator  120  to determine when the differential output signal OUTP-OUTN has transcended the voltage threshold represented by the differential reference voltage REIT-RUN. However, since the reference generator  110  is external, i.e., not an integral part of the detection circuitry  100 , its performance, in terms of consistency of the reference voltage, will vary with PVT in a manner substantially unrelated to the effects the PVT variations on the differential output signal OUTP-OUTN. Accordingly, the trip point of the voltage comparator  120  will also vary with PVT variations. 
         [0024]    Referring to  FIG. 5 , in accordance with one embodiment of the presently claimed invention, detection circuitry  200   a  implements an integrated reference generator in the form of a resistance RR across which an internal reference voltage VRES is generated in accordance with the current I provided by the current source  102  biasing the differential amplifier transistors Q 10 , Q 11 . The positive phase OUTP of the differential output signal is taken from between this resistance RR and its current source  102 , while the negative phase OUTN is taken from the emitter electrode of the common mode transistor Q 12 . As a result, when VID&lt;VREF and OUTN&gt;OUTP, the output  221  of the comparator  220  is low, and when VID&gt;=VREF and OUTP&gt;=OUTN, the comparator trips and the output  221  is high. 
         [0025]    This addition of an internal reference generator adds an offset to the output voltage, and advantageously eliminates the need for an external reference generator  110  and four-input voltage comparator  120  ( FIG. 4 ). However, compensation to ensure that the reference voltage VREF tracks the input signal amplitude VID over PVT variations would still be desirable. 
         [0026]    Referring to  FIG. 6 , compensation can be provided. As discussed above, an input stage  150  provides a buffered differential signal as the input signal for rectification by a rectification stage  200 , which, in turn, provides the differential signal to the voltage comparator  220 . An exemplary embodiment  150   a  as the input stage  150  includes N-type MOSFETs M 1 , M 2  mutually connected as a differential amplifier biased by a tail current source  152  with load resistances R. Connecting two additional resistances RC between the output electrodes (where such resistances RC are significantly higher than the load resistances R) and tapping the voltage between such resistances RC provides the common mode voltage of the differential output voltage BUFOUTP-BUFOUTN at the output electrodes of the input stage  150   a.  This common mode voltage VCM controls the common mode transistor Q 12  in the signal detection circuitry  200   b.    
         [0027]    As expressed by the equations, the input stage  150   a  has a signal gain equal to the product of the load resistance R and the transconductance GM of its transistors M 1 , M 2 . As is well known, transconductance GM and resistance vary over PVT. The internal reference voltage of the detection circuitry  200   b  should track these variations to minimize variations in the trip point over PVT. Accordingly, the current sources  102   a,    104   a  used to provide the biasing current I for the amplifier transistors Q 10 , Q 11  and common mode transistor Q 12  are implemented using transistors M 1 R, M 2 R that are replicas of the input stage transistors M 1 , M 2 . This ensures that the transconductance GM in the detection circuitry  200   b  tracks the transconductance GM in the input stage  150   a  over PVT variations. 
         [0028]    In accordance with an exemplary embodiment, the biasing voltages V 1 , V 2  for these replica transistors M 1 R, M 2 R, their biasing currents I and reference resistances RR are designed such that when the input voltage VII) is zero, the output voltage VOUT is less than the offset introduced by the internal reference voltage generator, and the comparator output  221  will be low. When the input voltage VIN to the input stage  150   a  reaches a value equal to the internal reference voltage, the output voltage VOUT (=OUTP−OUTN) becomes equal to zero and the comparator  220  trips, causing its output  221  to go high. 
         [0029]    In summary, the input stage  150   a  is a differential amplifier with an NMOS transistor pair that provides a nominal gain of two. Accordingly, the output signal has an amplitude double that of the input signal VIN, i.e., bufoutp−bufoutn=2*VIN. The output signal bufoutp-bufoutn of this stage  150   a  is provided to rectification circuitry Q 10 ,Q 11 ,Q 12 ,C within the detection circuitry  200   b.  The resulting signal has a magnitude half of that of the input signal, i.e., bufoutp−bufoutn)/2. The capacitor C holds the voltage at its peak amplitude. So the final output efoutp-efoutn is 2*(VIN/2)=VIN. 
         [0030]    Referring to  FIG. 7 , detection circuitry  200   c  in accordance with another embodiment of the presently claimed invention includes additional replica transistors M 3 R, M 4 R connected in parallel with the serially connected reference resistances RR and replica current source transistors M 1 R, M 2 R. These additional transistors M 3 R, M 4 R have their gate electrodes biased by the biasing voltage of the opposite circuit branch. For example, the biasing current source transistor M 1 R of the amplifier circuit branch is biased by a first biasing voltage V 1  while the current source transistor M 2 R of the voltage follower circuit branch is biased by a second biasing voltage V 2 . Meanwhile, the added transistor M 3 R in the amplifier circuit branch is also biased by the second biasing voltage V 2 , while the added transistor M 4 R across the voltage follower circuit branch is also biased by the first biasing voltage V 1 . As will be readily appreciated and indicated in the schematic diagram, this is to ensure that the amplifier Q 10 , Q 11  and common mode Q 12  transistors have the same total biasing current 11+12. 
         [0031]    These biasing voltages V 1 , V 2  can be provided with a reference voltage bias circuit  160  formed by the serial connection of equal resistances R 1 , R 2  and another replica transistor MR, biased by current provided from a current source  162 , with the gate electrode of the transistor MR biased by the voltage between the reference resistances R 1 , R 2 . The voltage across the upper resistance R 1  provides the first biasing voltage V 1 , while the voltage across the lower resistance R 2  provides the second biasing voltage V 2 . (In accordance with an exemplary embodiment, the biasing current is  100  microamps and the resistances R 1 , R 2  are selected to provide biasing voltages of 50 millivolts when conducting such current.) As a result, the reference voltage is provided by a circuit having gain variations over PVT variations similar to those of the input amplifier  150   a  ( FIG. 6 ). 
         [0032]    Referring to  FIG. 8 , the input stage  150   a,  reference voltage bias stage  160  and detection circuitry  200   c  with the rectifier and integrated reference voltage are interconnected in accordance with the identified inter-stage signals. 
         [0033]    Referring to  FIG. 9 , additional compensation can be provided by compensating for PVT variations in the base current of the common mode transistor Q 12 . As discussed above, the common mode output voltage VCM of the input stage  150   a  is sensed by a pair of resistances RC across the output of the input stage  150   a.  While these sensing resistances RC are significantly larger than the load resistances R so as to not load the output of the input stage  150   a,  there is nonetheless some amount of base current flowing through these resistances RC, i.e., base current for the common mode transistor Q 12 . This base current increases at low transistor gain (current gain β of transistor Q 12 ) and high temperature. This will introduce some variation in the sensed common mode voltage VCM over PVT variations and will translate to variation in the output voltage VOUT. This can be offset by using a replica circuit  170  in which a similar transistor Q 20  conducts an equal current I (=I1+I2), which is provided by replica transistors M 5 R, M 6 R biased by the biasing voltages V 1 , V 2 , similar to the detection circuitry  200   c.  The resulting base current of this replica common mode transistor Q 20  is mirrored by a current mirror circuit formed by transistors M 10  and M 11  to provide the base current needed by the original common mode transistor Q 12 . Accordingly, no base current for the common mode transistor Q 12  is drawn from input stage  150   a  via its sensing resistances RC. 
         [0034]    Referring to  FIG. 10 , integrated circuit (IC) design systems  204  (e.g., work stations or other forms of computers with digital processors) are known that create integrated circuits based on executable instructions stored on a computer readable medium  202 , e.g., including memory such as but not limited to CD-ROM, DVD-ROM, other forms of ROM, RAM, hard drives, distributed memory, or any other suitable computer readable medium. The instructions may be represented by any programming language, including without limitation hardware descriptor language (HDL) or other suitable programming languages. The computer readable medium contains the executable instructions (e.g., computer code) that, when executed by the IC design system  204 , cause an IC fabrication system  206  to produce an IC  208  that includes the devices or circuitry as set forth herein. Accordingly, the devices or circuits described herein may be produced as ICs  208  by such IC design systems  204  executing such instructions. 
         [0035]    Various other modifications and alternations in the structure and method of operation of this invention will be apparent to those skilled in the art without departing from the scope and the spirit of the invention. Although the invention has been described in connection with specific preferred embodiments, it should be understood that the invention as claimed should not be unduly limited to such specific embodiments. It is intended that the following claims define the scope of the present invention and that structures and methods within the scope of these claims and their equivalents be covered thereby.