Abstract:
A pulse shaper has a periodically reversible capacitor for pulse shaping. In performing testing of electronic components, the test specimens must be driven with different or, respectively, variable pulses. To this end, the pulse shaper has two mutually symmetrical circuits serving for the periodic charging and discharging of the capacitor, the circuits respectively containing binarilly switched adjustable resistors for setting the rise time and fall time of the pulse edges of the pulses to be formed, as well as containing a current source and a voltage source for the proportional change of the edge steepness with respect to the change of the shift of the pulses. In addition, voltage limitation is provided to generate the desired amplitude shift with respect to the baseline voltage with structure assigned to each circuit, the bias voltage sources of such structure being variable as a function of the shift to be set. The charging and discharging occurs by way of a periodically switched differential switch respectively comprising two transistors. The pulse shaper of the invention is particularly suited for automatic testing units for data processing devices.

Description:
This is a continuation of application Ser. No. 204,706, filed Nov. 6, 1980, now abandoned. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a pulse shaper which has a periodically reversible capacitor for the pulse shaping function. 
     2. Description of the Prior Art 
     In performance testing of electronic components, the test specimens must be driven with different or, respectively, variable pulses. It is therefore necessary to incorporate pulse shapers in the corresponding automatic testing units, the pulse shapers having output pulses which can be adapted to the required individual test conditions as close to practice as possible and in a programmable manner. In known pulse shapers, the edge rise times change given programming of shift and base voltages. 
     SUMMARY OF THE INVENTION 
     The object of the present invention is to provide a pulse shaper for generating pre-programmable pulses whose parameters are freely selectable and can be adjusted non-reactive with respect to one another. 
     In order to achieve the above object, a pulse shaper constructed in accordance with the present invention is designed in such a manner that two mutually symmetrical circuits, serving for the periodic charge and discharge, are provided, the symmetrical circuits respectively containing binarilly switched, adjustable resistors for setting the rise or, respectively, fall times of the pulse edges of the pulses to be formed, as well as containing a current source and a voltage source for the proportional change of the edge steepness with the change of the shift of the pulses. Moreover, a structure for voltage limitation which generates the desired shift is assigned to each circuit, the bias voltage sources of such structure being variable as a function of the shift to be generated. The charge and discharge occurs via periodically-operated differential switches respectively comprising two transistors. 
     By the above features, one obtains a pulse circuit in which the generated pulses can be adjusted at random both as to their base voltages as well as to their shift and with respect to their rise and fall times at the leading and trailing edges. 
     Further, it is advantageous that the binarilly switched resistors are realized by digital/analog converters which evaluate the data parameters input via a data bus and memory for the chronological rise and fall of the pulse edges. The shift and base voltages, which are likewise supplied via bus lines to a respective digital/analog converter via a memory, are loaded at their output side with ohmic resistors at which voltages proportional to the output currents arise, the voltages respectively serving as control voltages for the non-inverting inputs of two first operational amplifiers, below referenced OP1 and OP8. 
     By employing analog computers, a simple programmability is guaranteed for the pulse shape required for the respective case with simple manipulation. 
     Further, the invention can be described in such a manner that the operational amplifier assigned to the digital/analog converter for the shift voltage, together with a transistor connected at its output, forms a voltage source; and that an ohmic resistor is connected between the output of this source and the output of the operational amplifier post-connected to the digital/analog converter for the pulse base voltage. 
     By so doing, it is achieved that the programmed shift voltage always remains constant relative to the base voltage setting. 
     For the purpose of adding the shift voltage to the base voltage, the output of the current source post-connected to the digital/analog converter for the shift voltage has the control input of a further operational amplifier connected thereto, the inverting input of the further operational amplifier being connected to its output and its output being connected to a diagonal point of a Wheatstone bridge, whereas the output of the operational amplifier for the base voltage is fed to the other diagonal point of the Wheatstone bridge. 
     In order to always have symmetrical voltages which change in the same tendency available for the individual controls in the charge and discharge circuit, the Wheatstone bridge is constructed with a voltage divider in each of its branches. 
     Advantageously, the voltage sources in the charge and discharge circuits are designed as emitter followers which are formed by third operational amplifiers having respective transistors connected to their outputs, their control inputs being respectively connected to the tap of the voltage dividers located in the bridge branches lying obliquely opposite. 
     It is likewise advantageous that the current sources are formed by fourth operational amplifiers with respective transistors connected to their outputs and that the control inputs of these operational amplifiers are connected to the taps of the voltage divider of those further remaining bridge branches of the Wheatstone bridges lying obliquely opposite. 
     A further expedient embodiment provides that the limiter elements are Zener diodes and that their voltage sources are likewise fifth operational amplifiers having transistors connected to their outputs as emitter followers. The control inputs of these operational amplifiers are connected to the tap of the voltage dividers which lie between the emitter of the transistors of the current sources and a respective point of the bridge diagonals of the Wheatstone bridge. 
     The differential switches can be designed in such a manner that the two transistors forming a differential switch are respectively of the same conductivity, but nonetheless, of the complementary conductivity type with respect to the switches. The collectors of one respective transistor of each switch and the emitters of the two transistors belonging to one switch are connected to one another. The collectors of those transistors which still remain are directly connected to their voltage sources while by-passing the limiter elements. 
     By employing the above techniques, non-linear edge forms are avoided. 
     The memories connected to the inputs of the digital/analog converters can be designed as flip-flop memories. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Other objects, features and advantages of the invention, its organization, construction and operation will be best understood from the following detailed description, taken in conjunction with the accompanying drawings, on which: 
     FIG. 1 is a simplified block diagram of a pulse shaper constructed in accordance with the present invention; 
     FIG. 2 is a complete schematic diagram of the pulse shaper of FIG. 1; and 
     FIGS. 3 and 4 together form a logic portion for the pulse shaper. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     A simplified block diagram is illustrated in FIG. 1 in which the structure and the operation of the pulse shaper are set forth. The data for the baseline pulse voltage, the voltage amplitude, and the rise and fall times for the leading and trailing edges of the pulses arrive over a data bus DAB at preparation inputs of flip-flop memories 3, 4, 7 and 8. Four mutually-independent transfer clocks UT load the data into the corresponding flip-flop memories. The outputs of the flip-flop memories 3, 4, 7 and 8 respectively drive a digital/analog converter 1, 2, 5 and 6, whereby the digital/analog converters 1 and 2, for generating the amplitude and the base line pulse voltage generate respective binarily-graduated constant currents at their outputs and the digital/analog converters 5 and 6 determine the rise and fall times for the leading and trailing edges of the pulse. 
     With the assistance of a resistor R169, a current-proportional voltage amplitude is generated at the output of the digital/analog converter 2, the voltage amplitude being supplied to an operational amplifier OP8. The base line voltage is emitted at the output of the operational amplifier OP8, which serves as an impedance converter. 
     A current-proportional voltage likewise arises at a resistor R77 which is disposed at the output of the digital/analog converter 1, the voltage controlling a constant current source comprising an operational amplifier OP1 and a transistor T25. 
     The constant current generated at the collector of the transistor T25 flows through a resistor R82. Since the resistor R82 is connected at one side with the line for the base line voltage, its voltage drop is added, floating with respect to the base line voltage. An operational amplifier OP6, which is likewise connected as an impedance converter, senses this voltage and forms the amplitude voltage therefrom. By way of this circuit arrangement it is achieved that the programmed amplitude voltage always remains constant relative to the baseline voltage, i.e. when the baseline voltage changes, then the amplitude voltage also changes absolutely by the same amount. 
     By way of a pair of Zener diodes ZD4/DZ3 or ZD2/ZD1, the base line voltage is applied to a Wheatstone bridge 9 comprising a plurality of resistors R51, R52/R53, R54 and R144, R145/R143, R146. 
     In FIGS. 1 and 2, the voltage in the diagonal branch of the Wheatstone bridge 9 is tapped across a pair of operational amplifiers OP12/OP10 or, respectively, OP5/OP2, whereby the operational amplifier OP12 together with a transistor T65 and a operational amplifier OP2 together with the transistor T23 respectively form a constant voltage source, the operational amplifier OP10 with a transistor T71 and the operational amplifier OP5 together with a transistor T24, by contrast, operates as a constant current source by way of resistors of digital/analog converters 5, 6 for the leading and trailing edges of the pulses. It is achieved by the above measures that the constant current at the collectors of the transistors T24 or, respectively, T71 change only with respect to the shift voltage U HUB , but are not influenced by the variation of the base voltage U BASELINE . This is advantageous because, given variation of the shift voltage, the edge current should behave proportionally to the shift voltage according to 
     
         J.sub.FL =(U.sub.Hub ·C.sub.T Konst) 
    
     where t F1  is the Edge Duration, C T  is the capacitance of the capacitor C T  and J FL  is the charge current, if one wishes to obtain a shift-independent edge duration of the pulse. The basic value of the current-defining resistance R VFL  /R RFL  therefore, is set by the digital/analog converters 5, 6; the shift-dependent current control occurs by way of the diagonal bridge voltages of the Wheatstone bridge 9 to be described in greater detail below. Pairs of transistors T67/T68 and T27/T28 form two differential current switches which are respectively supplied on their emitter side by way of the transistors T71 and T24 with the shift-dependent constant current. 
     The transistors T68 and T28 are alternately driven via a trigger pulse AJ, so that current flows either across the transistors T68 or across the transistor T28. This current charges a capacitor C T  until one of two amplitude limiting diodes D12 or D34 becomes conductive and diverts the current towards constant voltage sources comprising components T70/OP9 or, respectively, T30/OP7. The above constant voltage sources are controlled across the resistors R89 and R148 directly by the pulse base voltage U BASELINE  or, respectively, shift voltage U HUB . Further, the error of the amplitude limiter voltage caused by the parasitic forward voltage of the limiter diodes D12 and D34 is identified by way of pairs of diodes D33 and D11 and is compensated by the operational amplifiers OP7 or, respectively, OP9. 
     This is possible because, in the final analysis, the same current flows through the diodes D33 and D11 as flows through the limiter diodes D12 and D34, the same forward voltage thus also occurs given diodes of the identical type. While by-passing the diodes D12 and D34, the transistors T67 and T27 see to it that the constant voltage sources with the transistors T30 or, respectively, T70 are always loaded with the same current during the switching intervals as in the limiting case. By so doing, undesired voltage fades given load jumps are largely avoided due to the finite internal resistance at the constant voltage sources with the transistors T30 and T70. 
     The pulse shaper signal E1 which has been completely edited as to its pulse parameters and as it arrives at the input of the output stage, now lies at the capacitor C T . The output stage, for example, represents a bipolar voltage follower having the voltage transformation ratio 1:1. 
     It functions as an impedance converter. 
     When the trigger pulse is reversed (see FIG. 2), and when an input B58 is `1` and and input B59 is `0`, then there applies ##EQU1## B T .sbsb.1 =Gain of the transistor T 1  J C  (T 2 )=0, because the transistor T 2  is not conducting ##EQU2## B T .sbsb.43 =Gain of the transistor T 43  J C  (T 42 )=0 because the transistor T 42  is not conducting 
     By way of the voltage drop at a pair of resistors R83/R84 or, respectively, R132/R133, the above collector currents J C  supply the differential drive signal for the transistors T26, T27, T28, T29 or, respectively, T66, T67, T68, T69. The voltage drop at the resistors R47 and R136 forms the offset. 
     There then applies the relationships: 
     
         U.sub.B (T.sub.26)/U.sub.B (T.sub.27)=U.sub.E (T.sub.23)-/J.sub.c (T.sub.42) (R.sub.47 +R.sub.83)/ 
    
     
         U.sub.B (T.sub.28)/U.sub.B (T.sub.29)=U.sub.E (T.sub.23)-/J.sub.C (T.sub.43) (R.sub.47 +R.sub.84)/ 
    
     
         U.sub.B (T.sub.66)/U.sub.B (T.sub.67)=U.sub.E (T.sub.65)+/J.sub.C (T.sub.1) (R.sub.136 +R.sub.133)/ 
    
     
         U.sub.B (T.sub.68)/U.sub.B (T.sub.69)=U.sub.E (T.sub.65)+J.sub.c (T.sub.2) (R.sub.136 +R.sub.133)/ 
    
     U B  =Transistor Base voltage 
     U E  =Transistor Emitter voltage 
     The transistors T26 and T29 or, respectively, T66 and T69 respectively together form a differential current switch whose collector current is determined by the relationships: ##EQU3## 
     The collector currents J c  of the transistors T26 and (T66) cause a voltage drop 
     
         U.sub.R85 =J.sub.c T.sub.66 ·R.sub.85 or, respectively, 
    
     
         U.sub.R134 =J.sub.C (T.sub.26)·R.sub.134 
    
     at the resistors R85 and R134, the voltage drop serving to generate an additional, dynamic cut-off voltage (addition of the collector currents of the transistors T26/T27 or, respectively, T66/T67) at the diodes D12 and D34 in the case of transfer. 
     Were this measure not undertaken, then, in the limiting case, the respectively saturated, conductive limiting diode D12 or D34, given slow edge rise times (Δ small reversal currents for the capacitor C T ), would more quickly discharge the capacitor C T  in the first moment upon transfer than is desired by means of the programmed reversal current, which would lead to a non-linear edge shape. 
     The semiconductor combination T27/T28 and T67/T68 is respectively interconnected with the transistors T24 and T74 to form a cascade stage, so that the following collector voltage derives: 
     
         U.sub.c (T.sub.24)=U.sub.B (T.sub.27)+/U.sub.BE (T.sub.27)/ or U.sub.B T.sub.28 +/U.sub.BE T.sub.28 / 
    
     
         U.sub.C (T.sub.74)=U.sub.B (T.sub.67)-/U.sub.BE (T.sub.67)/ or U.sub.B (T.sub.68)-/U.sub.BE (T.sub.68)/ 
    
     The transistors T24 and T71 respectively form constant current sources whose collector currents are determined by the relationships: ##EQU4## Thereby: B T .sbsb.24 =Gain of the transistor T 24  ; 
     U OP2  3=Input voltage at the operational amplifier OP2 Pin 3; 
     U OP5  3=Input voltage at the operational amplifier OP5 Pin 3; and 
     R VFL  =Program resistance of the digital-to-analog converter 5. ##EQU5## Thereby: B T71  =Gain of the transistor T71; 
     U OP12  3=Input voltage at the operational amplifier OP12 Pin 3; 
     U OP10  3=Input voltage at the operational amplifier OP10 Pin 3; and 
     R RFL  =Programmed resistance of the digital-to-analog converter 6. 
     In the above equations, the differential voltages U OP2  3-U OP5  3 or, respectively, U OP12  3-U OP10  3 are respectively automatically followed independently of the shift, in contrast thereto, the programmable resistors R VFL  and R RFL  must be externally pre-set by the program as quasi-static basic values. 
     It is guaranteed by means of this re-adjustment that the edge current changes in the same measure relative to the shift voltage according to the relationship 
     
         t.sub.FL =(U.sub.HUB ·C.sub.T)/J.sub.FL 
    
     t FL  =edge duration; and 
     J FL  =edge current. 
     As already mentioned, the outputs of the digital/analog converter 5, 6 function as resistors for the leading edge or, respectively, trailing edge programming. Since the programming occurs over ten bits, each digital/analog converter, abbreviated as DAU, contains 10 binarilly-staggered resistance values 
     DAU 5: R 7  /R 11  /R 15  /R 19  /R 23  /R 27  /R 31  /R 35  /R 39  /R 43   
     DAU 6: R 94  /R 98  /R 102  /R 106  /R 110  /R 114  /R 118  /R 122  /R 126  /R 130 , 
     which are switched on by the transistors: 
     DAU 5: T 4  /T 6  /T 8  /T 10  /T 12  /T 14  /T 16  /T 18  /T 20  /T 22   
     DAU 6: T 45  /T 47  /T 49  /T 51  /T 53  /T 55  /T 57  /T 59  /T 61  /T 63 . 
     Further, each of the two digital/analog converters must be constructed floating because of the readjustment. This occurs by means of the transistors: 
     DAU 5: T 3  /T 5  /T 7  /T 9  /T 11  /T 13  /T 17  /T 19  /T 21   
     DAU 6: T 44  /T 46  /T 48  /T 50  /T 52  /T 54  /T 56  /T 58  /T 60  /T 62 . 
     which respectively function as power source switches and thus reshape the OV-related input level of the TTL drive logic into a current shift of constant magnitude with, for example, ##EQU6## where U tt1  `1` is the control voltage for a TTL circuit when the input is a binary &#34;1&#34;. 
     This current serves as the base current for the transistor T4, so that the transistor T4 becomes saturated and becomes conductive except for the residual voltage U CE . In the cut-off case, the resistors R4 and R6 clear the base charges in that they clamp the bases of the transistors T3 and T4 to emitter potential. 
     The level converters of the remaining digital/analog converter bits operate analogously. The digital/analog converters for the basic voltage 2 and for the shift voltage 1 exhibit a current source output whose output currents are binarilly weighted. Since the current source outputs must be loaded toward 0 volts, on the one hand, and, on the other hand, the currents must be converted into voltages with the necessary d.c. offset, it is necessary to insert a pair of operational amplifiers OP1 or, respectively, OP8, whose output voltage behaves as follows: 
     U AOP   1  =J ADAU  ·R 78  +(12·R 78 )/R 76   
      where J ADAU  is the output current from the converter 1, and ##EQU7## where I ADAU  is the output current from the converter 2. 
     The output voltage of the operational amplifier OP8 can be directly employed as the pulse basic voltage, since the operational amplifier OP8 simultaneously operates as an impedance converter and the output voltage is therefore sufficiently loadable. 
     The output voltage of the operational amplifier OP1 serves as the control voltage for the constant current source comprising the operational amplifier OP3 and the transistor T25, for whose output current there applies: ##EQU8## Thereby: U K  1=output voltage of a constant voltage source K1 
     U OP3  3=input voltage at the operational amplifier OP3 Pin 3 
     B T  25=gain of the transistor T 25   
     The collector current J c  of the transistor T25 flows through the resistor R82 which is in turn connected to the pulse basic voltage. The voltage at the input of an operational amplifier OP6 is thus determined by the expression 
     
         U.sub.OP6 3=U.sub.BASELINE +J.sub.C (T.sub.25).R82 
    
     The operational amplifier OP6 operates as a voltage follower and an impedance converter; its output generates the shift voltage. 
     The output voltage of the constant voltage source K1 can be slightly changed by a potentiometer P1, whereby there is a possibility of setting the d.c. offset of the shift voltage relative to the pulse basic voltage. Basic voltage and shift voltage control two Wheatstone bridges whose diagonal voltages behave as follows: ##EQU9## 
     The above voltages respectively serve as guidance voltages for the edge current sources. The effects of temperature of the components are compensated by the arrangement of the components in a bridge circuit since the guidance voltages arise differentially. 
     By way of the resistors R89 and R148, respectively, the pulse voltage and the shift voltage directly control a respective constant voltage source comprising the components OP7/T30 or, respectively, OP9/T70. These two voltage sources generate the limiting voltages for the amplitude limiting diodes D12 and D34. After the limiter current either flows across the resistor R85 and the diode D12, or across the resistor 134 and the diode D34, there occurs an error with respect to the limiting voltage of: 
     
         U.sub.FHubbegr. =J.sub.C (T.sub.28).R.sub.134 +/U.sub.D34 
    
     
         U.sub.FBasisbegr. =J.sub.C (T.sub.68).R.sub.85 +/U.sub.D12 
    
     This error is compensated in that the same error voltage drop is intentionally generated at the resistor/diode combinations R44/D11 or, respectively, R131/D33 (the same limiter current flows through these components), and is communicated in equiphase to the inputs of the operational amplifier OP9 or, respectively, OP7 by an inverse feedback, via the resistors R87/R88 or, respectively, R142/R149. The operational amplifiers thereby automatically adjust the respective limiting voltage by the amount of the error. The capacitors C4, C5, C6, C10, C11, C12, C13, C14 serve for frequency compensation; they prevent an oscillation given re-programming of the limiting voltages. 
     The guidance voltages for the power supply of the output stage are tapped behind the Zener diodes ZD2 and ZD4. 
     FIGS. 3 and 4 illustrate the digital portion 3, 4, 7, 8 of the pulse shaper. The parameter data arrive in parallel at the preparation inputs of the D-flip flops J7, J8, J10-J18 via the inputs A3-A12, B41, B46, B49-B54, B56, B57 and by way of the inverter gates J2, J3, J4, J19, J23, J24. 
     A transfer clock (at the inputs A14, A15, B47, B48) now controls ten repective parallel clodk inputs of the above D-flip-flops by a power inverter, the outputs of the D-flip-flops leading directly to the corresponding DAU&#39;s in the analog portion. So that the parameter data are properly accepted into the flip-flops, it is necessary that the respective transfer clock be no narrower than 20 ns and that a minimum preparation time of 20 ns of the parameter data with respect to the negative edge of the transfer clock not be fallen below. 
     The change-over of the output resistance occurs via a relay I driven by the flip-flop J20. Given simultaneous activation of ELEMBF21-N and RELRTAKT-N, the flip-flop J20 is set, which effects the output resistance ≦3Ω. Given simultaneous activation of ELEMBF22-N and RELRTAKT-N, the flip-flop J20 is reset, whereby the output resistance 100Ω is switched on. 
     The pulse shaper output is fundamentally switched on or off with the relay II via the flip-flop J20. For this purpose, either ELEMBF21-N and RELRTAKT-N must be activated, which means connection of the pulse shaper output, or ELEMBF24-N and RELRTAKT-N must be activated, which means disconnection of the pulse shaper output. 
     Both flip-flops J20 are reset independent of the programming by a reset signal RELRUECK1-P, i.e. the pulse shaper output resistance Δ100Ω and the pulse shaper output are switched off. 
     The output pulse is distributed to five mutually independent output pins A50, B11, A58, A55 and B10 by the relays III, IV, V, VI and VII. The above relays are controlled by the flip-flops J5 and J6, whereby the flip-flops J5 (D-flip-flop) accept the information of ELEMBF-21N-ELEMBF-24N via the transfer clock ZELRTAKT-N, in contrast whereto the flip-flops J6 is set or, respectively, reset by ELEMBF19-9, or respectively, ELEMBF20-N in coincidence to RELRTAKT-N. Further, the flip-flop J6 can also be reset independently of the programming via RELRUECK2-P. 
     Although I have described my invention by reference to particular illustrative embodiments thereof, many changes and modifications of the invention may become apparent to those skilled in the art without departing from the spirit and scope of the invention. I therefore intend to include within the patent warrented hereon all such changes and modifications as may reasonably and properly be included within the scope of my contribution to the art.