Abstract:
A current generator generates a non-linear output current whose temperature coefficient exhibits a prescribed non-linear-to-quasi-linear curvature when a control voltage range is restricted. This particular current characteristic enables a voltage reference employing the current generator for high-order curvature correction to produce an output voltage whose variation is extremely flat over its industry standard operational temperature range.

Description:
FIELD OF THE INVENTION 
     The present invention relates in general to temperature-compensated electronic reference circuits and components therefor, and is particularly directed to a new and improved voltage-controlled current generator, which is operative to generate an output current that exhibits a prescribed non-linear to linear characteristic with temperature when its control voltage range is restricted. Injecting this output current into a voltage reference circuit, such as a ‘Brokaw’ bandgap voltage reference, provides improved high-order curvature correction, yielding an output voltage whose variation over a temperature range (e.g., −20° C. to +125° C.) is extremely flat (e.g., within several hundreds of microvolts). 
     BACKGROUND OF THE INVENTION 
     FIG. 1 is a reduced complexity diagram of a conventional first-order, current-based bandgap voltage reference, which generates an output voltage that is substantially independent of temperature, by summing a plurality of components whose temperature coefficients vary in a mutually complementary manner. For this purpose, a current I 1  proportional to absolute temperature (PTAT) is supplied to a series circuit of a diode D 1  and a resistor R 1  (referenced to ground (GND)). The voltage V1 across diode D 1  has an inverse or complementary to absolute temperature (CTAT) characteristic. As a result, if the (PTAT) current I 1  is inversely proportional to the value of a resistor having the same temperature coefficient as that of the resistor R 1 , the temperature behaviors of the respective voltage drops across diode D 1  and resistor R 1  will be mutually complementary, making the output voltage V 2  at a terminal OUT substantially (to a first order) independent of temperature. 
     A non-limiting example of what is commonly referred to as a ‘Brokaw cell’ current mirror implementation of the temperature-compensated bandgap voltage reference of FIG. 1 is schematically shown in FIG. 2. A current mirror circuit is formed of a first pair of MOSFETs including a MOSFET M 1  and a diode-connected MOSFET M 2  in a current mirror first leg containing a diode-connected MOSFET M 5 , and a second pair of MOSFETs comprised of diode-connected MOSFET M 3  and MOSFET M 4  in a second current mirror leg containing a MOSFET M 6 . MOSFETs M 1  and M 4  have their source-drain paths coupled in series with those MOSFETs M 5  and M 6  between voltage rail VDD and GND. 
     Diode-connected MOSFET M 2  has its gate connected in common with the gate of MOSFET M 1 , while MOSFET M 4  has its gate connected in common with the gate of diode-connected MOSFET M 3 . MOSFET M 2  has its source-drain path coupled in series with the collector-emitter path of a bipolar NPN transistor Q 1  and resistors R 2  and R 3  to GND. In a complementary manner, MOSFET M 3  has its source-drain path coupled in series with the collector-emitter path of a bipolar NPN transistor Q 2  and resistor R 3  to GND. The bases of transistors Q 1  and Q 2  are coupled to a voltage output terminal OUT. A MOSFET M 7  has its source-drain path coupled between voltage rail VDD and output node OUT, to which an output resistor RL referenced to GND is coupled. MOSFET M 7  has its gate coupled to the drain of MOSFET M 4 . 
     In the current mirror-based implementation of FIG. 2, the current flowing through MOSFETs M 2  and M 3  corresponds to the base-emitter difference voltage ΔVBE divided by the value of resistor R 2 , and is PTAT. Thus, the current I 1  supplied through resistor R 3  produces a PTAT voltage thereacross which is combined with the CTAT VBE voltage V 1  across transistor Q 2  to derive an output voltage reference V 2  having a first-order compensated temperature coefficient. As shown in FIG. 3, the voltage V 2  of the bandgap reference circuit of FIG. 2 varies with temperature in a substantially parabolic manner, and has a total variation on the order of 6.7 mV. A first order voltage references of the type shown in FIG. 2 is capable of producing a reference voltage whose temperature coefficient typically falls between 20 to 100 ppm/° c. 
     FIG. 4 illustrates a high-order compensating modification of the current-based voltage reference of FIG. 1, which employs an additional current component I 2  having a non-linear temperature coefficient. This additional non-linear current is intended to compensate for high-order, temperature dependent terms from the contribution of voltage V 1 . In the voltage reference of FIG. 4, the resistor R 1  of reference circuit of FIG. 1 is shown as series-connected resistors R 4  and R 5 , with an additional, high-order compensation non-linear current I 2  being supplied to the common connection of these two resistors. 
     One example of this type of bandgap voltage reference circuit that injects an additional non-linear current is disclosed in the U.S. Pat. No. 6,157,245 to Rincòn-Mora. (For a non-limiting example of additional prior art documentation showing another type of current-based bandgap reference circuit, attention may be directed to the U.S. Pat. No. 5,952,873 to Rincòn-Mora.) The above-referenced &#39;245 patent describes that second-order compensation may be achieved by injecting an additional non-linear current whose temperature coefficient is proportional to the ‘square’ of PTAT (or I 2 PTAT), so that its characteristic is parabolic. 
     While the squared temperature coefficient of the additional current tends to provide a second-order improvement at relatively low temperatures, where the variation in slope of the parabolic (squared) temperature coefficient characteristic is relatively gradual, the performance of an I 2 PTAT current-based voltage reference undesirably degrades at higher temperatures, due to the increasingly steep slope of the injected current&#39;s parabolic characteristic at such temperatures. 
     SUMMARY OF THE INVENTION 
     In accordance with the present invention, shortcomings of conventional first- and second-order compensated voltage references, including those described above, are substantially reduced by injecting into a voltage reference circuit, such as a ‘Brokaw’ voltage reference, a high-order, compensation current derived from a voltage controlled, non-linear current generator. This non-linear current generator is configured to generate an output current whose temperature coefficient exhibits a prescribed non-linear-to-quasi-linear curvature when the input or control voltage range is restricted. As will be described, this particular current characteristic enables a voltage reference that incorporates such a non-linear current generator for high-order curvature correction to produce an output voltage whose variation over an operational temperature range (e.g., −20° C. to +125° C.) is extremely flat (e.g., within several hundreds of microvolts). The inclusion of the non-linear current generator described herein in a bandgap reference allows a simple, power and area efficient method to achieve a curvature corrected output voltage. 
     To this end, the non-linear current generator according to the invention comprises an input transistor, referenced to a first power supply rail and having its collector-emitter path coupled in series with a PN junction device, such as a diode-connected transistor, to series-connected resistors that are coupled to a second power supply rail. The control electrode or base of the input transistor is coupled to receive an input or ‘reference’ (control) voltage, whose value is restricted or maintained within an ‘optimum’ range, in accordance with the desired operational parameters of the diode-connected transistor. In particular, this control voltage is set to a value, such that, in the low temperature region of operational temperature range, the diode-connected bipolar transistor operates just below the non-linear transition or ‘knee’ of its non-linear transfer characteristic. An output transistor has its emitter coupled to the common connection of the series resistors and its base coupled in common with the base of the diode-connected transistor. The collector of the output transistor is coupled to a current mirror, which mirrors the non-linear collector current from the output transistor as the desired non-linear output current I NL . 
     At cold temperatures and with an input voltage such that the voltage drop across the base-emitter of the input and output transistors causes the resistance of each branch to be large, the output current is very small. With an increase in temperature, the characteristics of the bipolar junction transistor cause the resistance of each collector-emitter path to decrease in an exponential fashion. As a consequence, the voltage across a summation resistor increases in the same exponential fashion and so does the output current. As temperature increases, the resistance of the collector-emitter paths of the transistors of the two branches becomes comparable to the resistance of the summation resistor, allowing some of the voltage drop from the input voltage to ground to be applied across the summation resistor. 
     The resistance of the series resistor is set such that it becomes larger than the decreasing collector-emitter resistance of the diode-connected transistor, so that its branch resistance stops its exponential decrease and becomes dependent on the resistance of the resistor in series with the summation resistor. The effect of the resistance of the diode-connected transistor and the series resistor branch being dominated by the series resistor, and thus the output transistor resistance becoming comparatively smaller, is such that the base-emitter voltage of the output transistor begins to decrease with temperature. 
     With the decrease in the base-emitter voltage of the output transistor, its collector-emitter path resistance begins to increase again, until the effects of increasing temperature become more dominant again and cause the resistance to decrease. At temperatures above this point, the voltage across the summation resistor increases in proportion to the temperature coefficient of 2VBEs, which is on the order of (−1)(2)(−2 mV/° C.)=±4 mV/° C., on a first-order basis. 
     The characteristics of the output current of the non-linear current generator improve the temperature performance of the bandgap voltage reference. With a first-order bandgap voltage reference curve shifted toward colder temperatures, the added positive temperature coefficient of the non-linear current generator initially causes the decreasing output voltage to increase. Then, as the slope of the output current vs. temperature of the non-linear current generator begins to decrease, the output voltage starts to decrease, until the contribution of the non-linear current causes the output voltage to increase again. When the resistor values are properly chosen, an optimized output voltage temperature characteristic can be realized. 
     Summing the voltages across the series resistors and the base-emitter junction of the bipolar transistor whose base and emitter are coupled between the voltage output terminal and the series resistors causes the curvature-corrected reference voltage circuit, to which the non-linear current I NL  is injected, to produce a reference voltage whose variation with temperature is confined within a very narrow (several hundred microvolts) range, which corresponds to a relatively small temperature coefficient (on the order of 2 ppm/° C.). 
     In addition to being used as a source of high-order compensation current for a ‘Brokaw’ current mirror-based bandgap voltage reference, the non-linear current generator of the invention may be combined with other temperature-controlled current sources, to produce a high-order, temperature compensated output current reference I REF , which exhibits an output current vs. temperature variation, that is extremely narrow (e.g., on the order of only tens of nanoamps over a range of from −20° C. to 125° C.) 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a reduced complexity diagram of a conventional current-based bandgap voltage reference; 
     FIG. 2 is a schematic diagram of a ‘Brokaw’ current mirror-based implementation of the bandgap voltage reference of FIG. 1; 
     FIG. 3 shows the variation with temperature of the output voltage of the bandgap voltage reference circuit of FIG. 2; 
     FIG. 4 illustrates a higher order compensating modification of the current-based voltage reference of FIG. 1, which employs an additional current component having a non-linear temperature coefficient; 
     FIG. 5 is a circuit diagram of a voltage-controlled, non-linear current generator in accordance with the invention; 
     FIG. 5A shows a plurality of curves representative of the variation in collector current versus base-emitter voltage for different temperatures; 
     FIG. 5B shows a plurality of curves representative of the variation in collector-emitter path resistance versus base-emitter voltage for different temperatures; 
     FIG. 5C shows a plurality of curves representative of the variation in collector-emitter path resistance versus temperature for different values of base-emitter voltage; 
     FIG. 5D is a graphical plot showing the variation in base-emitter voltage VBE Q52  of transistor Q 52  of FIG. 5 with temperature; 
     FIG. 5E is a graphical plot showing the variation in collector current IC Q52  of transistor Q 52  of FIG. 5 with temperature; 
     FIG. 5F is a graphical plot showing the variation in collector current IC Q51  of transistor Q 51  of FIG. 5 with temperature; 
     FIG. 6 schematically illustrates the manner in which the current mirror-based voltage reference circuit of FIG. 2 may be modified to incorporate the voltage-controlled, non-linear current generator of FIG. 5; 
     FIG. 7 shows a non-linear variation with temperature of a voltage V NL  across the ground-coupled, current-summing resistor of the voltage reference of FIG. 6 due to the current supplied from the non-linear current generator of FIG. 5; 
     FIG. 8 shows the non-linear voltage variation V NL  of FIG. 7 riding on a linear V PTAT  voltage; 
     FIG. 9 shows a high-order temperature compensated bandgap voltage reference vs. temperature characteristic produced by the voltage reference of FIG. 6; 
     FIG. 10 shows a combination of the non-linear current generator of the invention with PTAT and CTAT current sources to produce high-order compensation reference current I REF ; and 
     FIG. 11 shows the variation of reference current I REF  with temperature curve of the composite circuit of FIG.  10 . 
    
    
     DETAILED DESCRIPTION 
     Attention is now directed to the circuit diagram of FIG. 5, which shows an embodiment of a voltage-controlled, non-linear current generator according to the present invention, that may be used to supply a high-order curvature correction current, and which is readily incorporatable into a ‘Brokaw’ type bandgap voltage reference shown in FIG. 2, described above. As pointed out above, and as illustrated in FIG. 5, the non-linear current generator of the present invention produces an output current I NL  (which is mirrored off a collector current I Q52C  of an output transistor Q 52  within a current output branch OB), with a positive temperature coefficient that varies non-linearly with temperature, when a control or input reference voltage V 5A  applied to an input transistor Q 53  in a current input branch IB is restricted or maintained within a prescribed range. 
     This prescribed control voltage range is such that, in the low temperature region of an operational temperature range, a PN junction device, shown as diode-connected (NPN) bipolar transistor Q 51 , installed within current input branch IB, operates just below the ‘knee’ of its non-linear I-V transfer characteristic. This serves to effectively ‘squeeze’ the voltage V R5B  across a current summing resistor R 5 B, which controls the magnitude of the output current I NL . As temperature increases, diode-connected transistor Q 51  operates over the knee portion of its transfer function, so as to provide a non-linearity whose shape provides the desired second-order correction. FIG. 5A shows the variation in collector current vs. base-emitter voltage for a plurality of different temperatures. 
     More particularly, in the non-linear current generator of FIG. 5 diode-connected transistor Q 51  has its collector-emitter current flow path connected in series with the collector-emitter current flow path of an input (NPN) transistor Q 53  and series-connected resistors R 5 A and R 5 B, between a pair of power supply rails VDD and GND. The base of input transistor Q 53  is coupled to receive an input or ‘reference’ (control) voltage V 5 A, whose value is restricted in accordance with the desired operational parameters of diode-connected transistor Q 51 , resistor R 5 A and transistor Q 52 , as described above. Output transistor Q 52  has its emitter coupled to the common connection of resistors R 5 A and R 5 B, and its base coupled in common with the base of the diode-connected transistor Q 51 . The collector of output transistor Q 52  is coupled to an input port  54  of a current mirror  55 , comprised of MOSFETs M 8  and M 9 , which mirrors the non-linear collector current from output transistor Q 52  at port  54  as non-linear output current I NL  at an output port  56 . 
     The non-linear current generator of FIG. 5 operates as follows. The parameters of the circuit are such that transistors Q 53  and Q 52  are biased in the forward active mode, while transistor Q 51 , being diode-connected, is forced to the edge of saturation. In addition, as pointed out above, the input voltage V 5 A applied to the base of input transistor Q 53  is such that it effectively restricts the voltage across R 5 B in the low temperature region of its operational temperature range. 
     As the temperature increases, the current through each current branch IB and OB, and therefore the voltage V R5B  across summation resistor  5 B, will increase in a non-linear (generally exponential) manner (over the ‘knee’ region of the diode transfer function of transistor Q 51 ), since the base-emitter voltages Vbe&#39;s of the transistors remain substantially constant, until Vbe reaches the forward base-emitter turn-on voltage Vbe ON . This means that the collector current I Q52C  of output transistor Q 52 , and therefore the output current I NL  at the output port  56  of the current mirror  55 , will behave in the same manner. 
     When the temperature reaches the Vbe ‘turn-on’ temperature, the branch currents (and therefore the output current I NL  of current mirror  55 ), as well as the voltage V R5B  across current summing resistor R 5 B, will transition from increasing in a non-linear (exponential) manner to a more linear fashion due to the dominant first order negative temperature coefficient of Vbe. 
     Namely, at cold temperatures and with an input voltage such that the voltage drop across the base-emitter of transistors Q 53  and Q 52  causes the resistance of each branch to be large (as shown in FIG.  5 B), the output current is very small. As the temperature increases, the characteristics of the bipolar junction transistor causes the resistance of each collector-emitter path (RCE) to decrease in an exponential fashion (as shown in FIG.  5 C). As a result, the voltage across resistor R 5 B increases in the same exponential fashion and so does the output current. With an increase in temperature, the resistance of the collector-emitter (RCE) paths of transistors Q 51  and Q 52  becomes comparable to the resistance of resistor R 5 B, allowing some of the voltage drop from voltage V 5 A to ground to be applied across resistor R 5 B. 
     The resistance of resistor R 5 A is set such that it becomes larger than the decreasing collector-emitter resistance of transistor Q 51 , so that its branch resistance stops its exponential decrease and becomes dependent on the resistance of resistor R 5 A. The effect of the resistance of the transistor Q 51 —resistor R 5 A branch being dominated by resistor R 5 A, and thus the transistor Q 52  branch resistance becoming comparatively smaller, is such that the base-emitter voltage VBE Q52  of transistor Q 52  begins to decrease with temperature (as shown in FIG.  5 D). 
     With the decrease in the base-emitter voltage of transistor Q 52 , the RCE of transistor Q 52  begins to increase again, until the effects of increasing temperature become more dominant again and cause the resistance to decrease. At temperatures above this point, the voltage across resistor R 5 B increases in proportion to the temperature coefficient of 2VBEs, which is approximately (−1)(2)(−2 mV/° C.)=±4 mV/° C., on a first-order basis (as shown in FIG. 5E; FIG. 5F shows the variation in collector current I Q51  of transistor Q 51  with temperature). 
     The characteristics of the output current of the non-linear current generator improve the temperature performance of the bandgap voltage reference. With a first-order bandgap voltage reference curve shifted toward colder temperatures (as shown in FIG.  3 ), the added positive temperature coefficient of the non-linear current generator initially causes the decreasing output voltage to increase (as shown in FIG.  9 ). Then, as the slope of the output current vs. temperature of the non-linear current generator begins to decrease, the output voltage starts to decrease, until the contribution of the non-linear current causes the output voltage to increase again. When the resistor values are properly chosen, an optimized output voltage temperature characteristic can be seen. 
     FIG. 6 schematically illustrates the manner in which the current mirror-based voltage reference circuit of FIG. 2 may be modified to incorporate the voltage-controlled, non-linear current generator of FIG.  5 . Components and connections of FIG. 6 that are identical to those shown in FIGS. 2 and 5 will not be redescribed. The augmented voltage reference circuit FIG. 6 differs from that of FIG. 2 by the addition of a resistor R 6 , coupled in series with resistor R 3  and referenced to ground, and coupling the non-linear current I NL  to the node between resistors R 3  and R 6 . Resistor R 6  corresponds to the resistor R 5  in the higher order compensation diagram of FIG. 4, described above. In addition, the circuit of FIG. 6 includes a start-up circuit  60  comprising a further bipolar (NPN) transistor Q 6  having its collector-emitter path coupled across the collector and base of transistor Q 1 , and its base coupled to a voltage divider circuit comprised of resistor R 7  and series-connected diodes D 2  and D 3  between VDD and GND. 
     FIG. 7 shows a non-linear variation  71  with temperature of the voltage V NL  across the ground-coupled resistor R 6  as a result of the non-linear current component I NL  injected through resistor R 6  from the current mirror  55  of the non-linear current generator. It can be seen that the non-linear portion of voltage curve  71 , which has a total variation on the order of only 20 millivolts over the entire temperature range, lies essentially at low temperatures and becomes relatively linear in the upper region of the temperature range (on the order of 75° C. and above) 
     FIG. 8 shows the non-linear voltage variation V NL  of FIG. 7 ‘riding on’ extending slightly upwardly from a linear V PTAT  voltage developed across resistors R 3  and R 6  due to the current I 1  flowing therethrough. Also shown in FIG. 8 is the CTAT voltage V 1 , corresponding to the base-emitter of transistor Q 2 , between output terminal OUT and the series-connected resistor pair R 3 -R 6 . Summing the voltage profiles V 1 +V PTAT +V NL  of FIG. 8 produces the high-order temperature compensated bandgap voltage reference vs. temperature characteristic of FIG. 9, which corresponds to that of the output voltage V 2  provided at the output terminal OUT. As shown therein, over an operational temperature range of −20° C. to +125° C., the output voltage V 2  is confined within a very narrow 360 microvolt range, which corresponds to a temperature coefficient of only 2.10 ppm/° C. 
     In addition to employing the non-linear current generator of the present invention as a source of high-order compensation current, as in the augmented voltage reference of FIG. 6, the non-linear current generator of FIG. 5, may be combined with other temperature controlled current sources, such as conventional complementary, temperature dependent (e.g. PTAT and CTAT) current sources, as diagrammatically illustrated in FIG. 10, to provide another embodiment of a high-order, temperature-compensated current reference I REF . In the embodiment of FIG. 10, the non-linear current source of FIG. 5, shown at  101 , is summed with currents produced by a PTAT current source  102  and a CTAT current source  103 , to produce a high-order, temperature-compensated output current reference I REF , which exhibits the output current vs. temperature curve of FIG.  11 . As shown therein, the range of variation of the resulting current reference I REF  is very narrow (only 50 nanoamps over a temperature range of −20° C. to 125° C., or only +/−1.24%). 
     While I have shown and described several embodiments in accordance with the present invention, it is to be understood that the same is not limited thereto but is susceptible to numerous changes and modifications as known to a person skilled in the art. I therefore do not wish to be limited to the details shown and described herein, but intend to cover all such changes and modifications as are obvious to one of ordinary skill in the art.