Abstract:
A switching power supply device has lower correction circuit losses, and enables adjustments without affecting overcurrent limiting or other characteristics. An integrated circuit IC for power supply control generates a switching signal based on a feedback signal from a feedback circuit and a voltage signal from a current detection input terminal, and outputs the switching signal from an output terminal to a switching element. A voltage controlled oscillator is provided which, when the load is judged to be light based on the magnitude of the feedback signal, lowers the switching frequency. The correction circuit is connected between the output terminal of the integrated circuit and the signal input terminal for current detection, acts only when the switching element is on, and has the function of further lowering the switching frequency set in the integrated circuit.

Description:
BACKGROUND OF THE INVENTION AND RELATED ART STATEMENT 
     The present invention relates to a switching power supply device which supplies prescribed output power to a load according to a preset voltage, and in particular relates to a switching power supply device which lowers the switching frequency of a switching element during light loading or no loading or standby (hereinafter also simply called “light loading”), thereby achieving reduced power consumption during light loading or no loading, or reduced standby power during standby. 
     In the prior art, IC circuits for switching power supply control have been utilized which improve power efficiency by lowering the switching frequency during-light loading, with the aim of lowering switching losses in the switching power supply (see for example Japanese Patent Laid-open No. 2007-215316 (page 1, line 15 to page 13, line 16, FIG. 4, FIG. 5)). In Japanese Patent Laid-open No. 2007-215316 (page 1, line 15 to page 13, line 16, FIG. 4, FIG. 5), a switching power supply control circuit is disclosed employing current mode control (positive detection method), in which the value of the current flowing in the power MOSFET (Metal-Oxide Semiconductor Field Effect Transistor) or other switching element is detected as a positive voltage signal. 
       FIG. 9  is a block diagram showing the control circuit of a quasi-resonant switching power supply disclosed in Japanese Patent Laid-open No. 2007-215316 (page 1, line 15 to page 13, line 16, FIG. 4, FIG. 5). The quasi-resonant switching power supply of  FIG. 9  is shown as only one example of a switching power supply of the prior art. This invention is not limited to quasi-resonant devices, and can also be applied broadly to switching power supplies other than quasi-resonant devices. 
     In the power supply control circuit  10 , a bottom detection circuit (valley detection)  11  is connected to the input terminal ZCD for zero current detection. The bottom detection circuit  11  is a comparator which compares the voltage applied to the input terminal ZCD with a reference voltage close to the 0 V level (threshold); the output terminal of this bottom detection circuit  11  is connected to one of the input terminals of the AND circuit  12 , and through the AND circuit  12 , is connected to the one-shot circuit  13 . The output terminal of the voltage controlled oscillator (VCO)  14  is connected to the other input terminal of the AND circuit  12 . The voltage controlled oscillator  14  is an oscillator which changes the output frequency depending on the magnitude of the input voltage (VCO voltage), and comprises a voltage signal input terminal VCO and a reset signal input terminal Reset. The VCO voltage input terminal of the voltage controlled oscillator  14  is connected to the input terminal FB for feedback signal VFB detection, and the input terminal for reset signals Reset is connected to the output terminal of the one-shot circuit  13 . 
     The input terminal FB for detection of the feedback signal VFB is connected to the inverting input terminal (−) of the comparator  15 . The non-inverting input terminal (+) of the comparator  15  is grounded via 0.5 V reference power supply E 1 , and a disable signal Disable is output from the output terminal to the inverter circuit  16 . The output of the inverter circuit  16  is connected to the clear terminal (CLR) of the one-shot circuit  13 . A 5 V reference power supply E 2  is connected via the series circuit of a resistor R and diode D to the input terminal FB, and determines the FB terminal voltage. 
     The current comparator  17  is connected to the signal input terminal for current detection IS, and a current detection signal is supplied to the non-inverting input terminal (+) among the four input terminals of the current comparator  17 . The remaining three inverting input terminals (−) are connected respectively to the input terminal for feedback signal VFB detection FB, the 1 V reference power supply E 3 , and the output terminal of the soft-start circuit  18 . The output terminal of the current comparator  17  is connected to the reset terminal R of the flip-flop circuit  19 . The set terminal S of the flip-flop circuit  19  is connected to the output terminal of the one-shot circuit  13 . The Q output terminal of the flip-flop circuit  19  is connected via the AND circuit  20  to the output terminal OUT, and the output signal Q of the flip-flop circuit  19  is output to the power MOSFET or other externally connected switching element Q 1  (see  FIG. 10  below) as a switching signal from the output terminal OUT. The soft-start circuit  18  generates a soft-start signal, which limits the turn-on interval of the switching element Q 1  during startup of the switching power supply. 
     The inverting input terminal (−) of the comparator for overload detection  21  is connected to the input terminal FB for feedback signal VFB detection, and the non-inverting input terminal (+) is grounded via the 3.3 V reference power supply E 4 . The output terminal of the comparator  21  is connected to the reset terminal Reset of the timer circuit  22 . The timer circuit  22  is used to set two delay times; the first output signal (low) is output to the AND circuit  20  100 ms after the comparator  21  detects an overload state, and forcibly halts the supply of the switching signal to the switching element Q 1 . 
     The second output signal of the timer circuit  22  is output 800 ms after an overload state is detected, and is supplied as a reset signal to a startup circuit, not shown, provided within the power supply control circuit  10 . 
     In this power supply control circuit  10  for a switching power supply disclosed in Japanese Patent Laid-open No. 2007-215316 (paragraphs [0002] to [0025], FIG. 4, FIG. 5), the voltage applied to the switching element Q 1  upon zero-cross detection is the minimum of the resonance waveform, and the switching element Q 1  is turned on with this timing to start the next switching cycle, in what is generally called a quasi-resonant type or partial-resonance type switching power supply control. 
     In the control circuit shown in  FIG. 9 , during normal operation the current signal of the switching element Q 1  is input to the input terminal IS, and the current comparator  17  compares this current signal with the feedback signal VFB input to the input terminal FB; the power supplied to the secondary side is controlled by making the current in the switching element Q 1  small when the load is light and making the current in the switching element Q 1  large when the load is heavy, executing control so that the output voltage is substantially equal to the voltage setting. 
     The feedback signal VFB input to the input terminal FB decreases when the load is light and the output voltage is high, and increases when the load is heavy and the output voltage declines. The voltage controlled oscillator  14  lowers the frequency more for a smaller feedback signal VFB, which is the VCO voltage, so that the lighter the load, the lower the oscillation frequency of the voltage controlled oscillator  14 , and the heavier the load, the higher the frequency. A detailed explanation is here omitted, but the frequency of the switching signal (the switching frequency) output from the output terminal OUT of the power supply control circuit  10  is governed by the oscillation frequency of the voltage controlled oscillator  14 , so that, in essence, the lighter the load, the lower is the switching frequency. This is because in light loading, the fraction of total losses accounted for by switching loss is increased, and so the frequency is lowered with the aim of alleviating the switching loss during light loading. This technique of lowering the switching frequency during light loading is also widely applied to switching power supplies other than quasi-resonant type devices. 
     The reference voltage E 3  (1 V) connected to the current comparator  17  is a reference voltage used to limit overcurrents in the switching element Q 1 . In the case of an overload and the like, the maximum value of the current signal is limited to the reference voltage E 3  (1 V) in order to protect the power supply circuit and the load. 
       FIG. 10  is a block diagram showing one example of a positive detection-type switching power supply device of the prior art. 
     The switching power supply device of  FIG. 10  supplies power from the primary-side DC input power supply V IN  of a transformer T 1  to the secondary-side load (not shown) according to a voltage setting. An LC resonance circuit is formed by the inductance (Lp) of the primary-side windings Lp of the transformer T 1  and the capacitance of the resonance capacitor Cr (which can also be only the parasitic capacitance of the switching element Q 1 ) connected in parallel with the power MOSFET or other switching element Q 1 . The input voltage V IN  is supplied to one end of the smoothing capacitor C 1  and to one end of the primary windings Lp of the transformer T 1 ; the other end of the primary windings Lp is connected to the drain of the switching element Q 1 . The source of the switching element Q 1  is connected, via the sense resistor Rs, to the other end of the smoothing capacitor C 1 , and the gate is connected via the resistance R 1  to the output terminal OUT of the integrated circuit IC. 
     The integrated circuit IC in the switching power supply circuit of  FIG. 10  is for example equivalent to the power supply control circuit  10  of  FIG. 9 ; in  FIG. 9 , only the zero current detection input terminal ZCD, feedback signal detection input terminal FB, signal input terminal for current detection IS, and the output terminal OUT for output of the control signal to the switching element Q 1  are shown. 
     The primary windings Lp, secondary windings Ls, and auxiliary windings Lb of the transformer T 1  are all wound around the same core of the transformer T 1 . The inductance of the secondary windings Ls is Ls, and the inductance of the auxiliary windings Lb is Lb. The resonance capacitor Cr is connected in parallel with the series circuit of the switching element Q 1  and the sense resistor Rs, but may be connected in parallel with the primary windings Lp. The auxiliary windings Lb are connected to a rectifying diode D 2  and smoothing capacitor C 2  which form the power supply of the integrated circuit IC. The resistor R 2  supplies the voltage at the connection point between the switching element Q 1  and the sense resistor Rs to the signal input terminal for current detection IS; and the resistor R 3  is provided to input the voltage across the auxiliary windings Lb as-is, without rectification, to the input terminal ZCD of the integrated circuit IC. The sense resistor Rs functions as a current detection element. 
     A diode D 3  and smoothing capacitor C 3  which rectify the voltage appearing across the secondary windings Ls are provided at the secondary windings Ls of the transformer T 1 . The anode of the diode D 3  is connected to one end of the secondary windings Ls, and the cathode is connected to the power supply output terminal Vout as well as to one end of the smoothing capacitor C 3 . The other end of the smoothing capacitor C 3  is connected to the other end of the secondary windings Ls, as well as to the ground terminal Gnd. 
     The level at the output terminal OUT of the integrated circuit IC changes between high and low to drive the gate of the switching element Q 1 , turning the switching element Q 1  on and off, and by this means the desired smoothed DC voltage is generated on the side of the secondary windings Ls of the transformer T 1 , between the power supply output terminal Vout and the ground terminal Gnd. During on intervals, a drain current flows in the switching element Q 1 , and a current flows on the side of the primary windings Lp of the transformer T 1  connected thereto, accumulating energy. Thereafter the switching element Q 1  turns off, but by means of the energy accumulated in the transformer T 1 , a current passes through the diode D 3  and flows in the smoothing capacitor C 3  on the side of the secondary windings Ls of the transformer T 1  during off intervals of the switching element Q 1 . In this way, a smoothed DC voltage appears on the side of the secondary windings Ls of the transformer T 1 , between the power supply output terminal Vout and the ground terminal Gnd. 
     Between the power supply output terminal Vout and the ground terminal Gnd is provided an output detection circuit comprising a series circuit of resistors R 5  and R 6 , a resistor R 7 , a photodiode PD comprising a phototransistor PT and a photocoupler, a capacitor C 4 , and a shunt regulator D 4 . Here, a current flows in the photodiode PD according to the output voltage (the current which flows is larger to the extent that the output voltage is higher than the voltage setting), the photodiode PD emits light in a quantity corresponding to this current, and a feedback signal is supplied to the phototransistor PT connected between the feedback signal detection input terminal FB of the integrated circuit IC and the ground terminal Gnd. The larger the quantity of light emitted by the photodiode PD, the larger the current which flows in the phototransistor PT, and this current flows in the resistor R, so that the voltage drop across the resistor R increases. That is, the higher the output voltage, the larger is the current flowing in the phototransistor PT, so that the feedback signal VFB becomes small. By means of this feedback function, a switching power supply device can supply power according to fluctuations in the load, not shown. The feedback circuit  25  comprises the portions enclosed within the dashed line. 
     The positive detection switching power supply device shown in  FIG. 10  has a sense resistor Rs as a current detection element, and is characterized in comprising an overload protection (OLP; also called overcurrent protection, OCP) function, which protects the load from excessive currents by means of an overcurrent limiting circuit which employs the signal obtained by applying a bias, by means of the resistors R 4  and R 2 , to the current detection signal (the signal itself being a voltage) detected by the sense resistor Rs. In recent devices, reduced power consumption in the power supply control circuit  10  itself has been sought, and a method of reducing the current flowing in the path from the input power supply V IN  through the resistors R 4 , R 2 , Rs is conceivable. Before explaining this method, first the function of the resistors R 4  and R 2  is explained. 
     First, a state in which the resistors R 4 , R 2  are not provided is considered. This overcurrent limiting circuit does not directly monitor overcurrent on the secondary side of the transformer T 1 , but instead monitors current changes on the side of the primary windings Lp to detect overcurrents to the load and halt switching operation. This is because when directly monitoring the secondary-side load current, a circuit to feed back a signal to the primary side becomes necessary. Specifically, the voltage across the sense resistor Rs which is the current detection element in  FIG. 10  is compared with a reference voltage which is an overload protection (OLP) judgment reference (and is hereinafter called the judgment reference voltage Vth), and when the voltage across the sense resistor Rs reaches this reference voltage, it is judged that an overcurrent has occurred. 
       FIGS. 11(A) and 11(B)  show the primary-side current waveforms corresponding to different input voltages. Here, when the respective input voltages V IN  are applied as V 1  and V 2 , the positive-detection current detection signal occurring across the resistor Rs is shown as the current waveform corresponding to the inductor current I L  flowing in the primary windings Lp of the transformer T 1 . 
     An inductor current I L  begins to flow in the primary windings Lp at the time that the switching element Q 1 , an N-channel MOS transistor, is turned on, and this current increases with a slope proportional to the input voltage V IN  (dI L /dt=V IN /Lp). When the current detection signal reaches the judgment reference voltage Vth for overload protection (OLP), the power supply control circuit  10  (integrated circuit IC) of  FIG. 9  judges that an overcurrent is occurring, and the switching element Q 1  is turned off. 
     In the positive-detection type switching power supply device shown in  FIG. 10 , a response lag time Δt occurs between the time an overcurrent is actually judged to have occurred in the integrated circuit IC and the time the switching element Q 1  is turned off. For this reason, as shown in  FIGS. 11(A) and 11(B) , overshoot exceeding the judgment reference occurs in the inductor current I L  actually flowing in the switching element Q 1  at the time of overcurrent limiting operation. While the slope of the inductor current I L  is proportional to the input voltage V IN , the response lag time Δt, which is determined by operation of the control system, is regulated by the power supply voltage of the power supply control circuit  10  (the integrated circuit IC in  FIG. 10 ), so that it is not affected by the input voltage V IN . Hence upon comparing the current detection signals from the sense resistor Rs for a case in which the input voltage V IN  is a small value (V 1 ) as shown in  FIG. 11(A) , and for a case in which the value is V 2  (&gt;V 1 ) shown in  FIG. 11(B) , the above-described overshoot amount ΔV is larger for higher values of the input voltage V IN  (ΔV 1 &lt;ΔV 2 ). 
       FIG. 12  shows changes in the inductor current during overcurrent limiting operation in the switching power supply device of  FIG. 10 . When the switching element Q 1  is cut off after a lag time, that is, during overcurrent limiting operation, the inductor current I L  flowing in the primary windings Lp of the transformer T 1  increases in proportion to the input voltage V IN , as explained in  FIGS. 11(A) and 11(B) . In a conventional positive-detection type switching power supply device, when for example the device is used as the power supply for a personal computer in Japan, the 100 V commercial AC power supply is rectified and smoothed for use as the DC input power supply. In other countries, a 200 V AC power supply may be used. On the other hand, only voltages of at most approximately 10 to 20 V are required as the output voltages from the secondary windings Ls or from the auxiliary windings Lb of the transformer T 1 . When there is deviation in the voltage of the commercial AC power supply, that is, in the input voltage V IN , if the higher the input voltage V IN  the larger the inductor current I L  which flows when turning off the switching element Q 1 , problems are posed for power supply safety. 
     Hence with the aim of correcting for this overshooting in the switching power supply device shown in  FIG. 10 , a resistance circuit in which resistors R 2  and R 4  are connected in series is provided. By means of this resistance circuit, the voltage level of the sense resistor Rs is shifted in the positive direction. The amount of level shifting is larger for higher input voltages V IN , so that the higher the input voltage V IN , the more quickly an overcurrent state can be judged in the stage before the voltage of the sense resistor Rs reaches the overcurrent limiting judgment reference voltage Vth. Hence the overshoot amount ΔV when actually turning off the switching element Q 1  can be compensated by this resistance circuit. 
     However, in a positive detection method in which the level is shifted by a resistance circuit, when considered from the standpoint of reduction of power consumption under light loading or no loading or reduction of standby power during standby, which has emerged as an issue in power supply systems in recent years, the power consumption due to the current flowing from the input power supply V IN  (in a normal power supply system, the input power supply V IN  is at the highest voltage) through the resistance circuit of the resistors R 4 , R 2 , Rs to ground (Gnd) becomes a problem. Hence a method is known in which, in order not to pass unnecessary current in a switching power supply device, while compensating for the phenomenon in which the higher the input voltage V IN , the larger is the overcurrent when the switching element Q 1  is turned off, as shown in  FIG. 12 , the current detection signal level is shifted in the negative direction, to achieve reduced power consumption (see for example Japanese Patent Laid-open No. 2003-299351 (page 8, line 2 to page 9, line 24, FIG. 2)). 
       FIG. 13  is a block diagram showing one example of a negative-detection type switching power supply device of the prior art. 
     As for example in the case of the switching power supply device disclosed in Japanese Patent Laid-open No. 2003-299351 (page 8, line 2 to page 9, line 24, FIG. 2), a negative-detection type switching power supply device is configured such that current detection means employs a sense resistance Rs to detect the current flowing in the primary windings, or the current flowing in the switching element, as a negative voltage. Hence in the switching power supply device shown in  FIG. 13 , the signal input terminal for current detection IS and the sense resistor Rs are connected via a resistor Ra. Moreover, the signal input terminal IS is connected to the connection point between the auxiliary windings Lb and the rectifying diode D 2 , and to the power supply terminal VCC supplying power to the integrated circuit IC, via the resistor Rb and the correction resistor Rc, respectively. 
     First, the functions of the resistors Ra and Rb are explained (the function of the correction resistor Rc is explained below). As is clear from the circuit configuration in  FIG. 13 , the larger the current on the primary side, the larger the absolute value of the negative voltage which becomes the current detection signal. The resistors Ra and Rb correspond to the respective resistors R 2  and R 4  in the positive detection method of  FIG. 10 , and are provided to apply a negative bias to the current detection signal. During intervals in which the switching element Q 1  is turned on, a negative potential appears at the connection point between the auxiliary windings Lb and the rectifying diode D 2 . This negative potential is insulated from the smoothing capacitor C 2  by the rectifying diode D 2 , and so is proportional to the input voltage V IN  (but with the sign inverted). Hence just as when the positive-voltage current detection signal has a positive bias applied in the positive detection method, so in the negative detection method the negative-voltage current detection signal has a negative bias applied, which is proportional to the input voltage V IN . 
     In the positive detection method and the negative detection method, the power consumed in the resistors R 4 , R 2 , Rs and in the resistors Rb, Ra, Rs differs greatly. This is because the power consumed in a resistor is proportional to the square of the voltage applied to the resistor ((voltage) 2 /resistance value), and the applied voltages differ greatly. As explained above, when a commercial AC power supply is rectified and smoothed to obtain an input voltage V IN , the value is approximately 100 to 200 V, whereas the output voltage (absolute value) from the auxiliary windings Lb is at most approximately 10 to 20 V, so that the power consumption can be reduced by about two orders of magnitude. 
     In the power supply control circuit  10  (IC circuit) of  FIG. 13 , only a portion of the elements comprised by the circuit is shown. Here, the voltage controlled oscillator  14 , current comparator  17 , and flip-flop circuit  19  are circuits corresponding to the control circuit shown in  FIG. 9 , a signal inversion circuit  23  is positioned to supply signals to the non-inverting input terminal (+) of the current comparator  17  from the feedback signal VFB detection input terminal FB, and a level shift circuit  24  is provided between the current detection input terminal IS and the inverting input terminal (−) of the current comparator  17 . Although omitted in  FIG. 13 , the power supply control circuit  10  also comprises a zero current detection input terminal ZCD, a terminal VH to which startup current is supplied, and the like. 
     The voltage controlled oscillator  14  is an oscillator used to determine the switching frequency; and the oscillation frequency is controlled by a feedback signal VFB (this signal is equivalent to a so-called error signal), output from the feedback circuit  25 , resulting from amplification of the difference between the voltage output to the load and the voltage setting. The frequency characteristic is such that, in the range in which the load is judged to be light (for example, when the feedback signal VFB is 0.9 V or less), the frequency is proportional to the voltage of the feedback signal VFB, and declines substantially linearly to the minimum frequency. When the load is heavy, the frequency is constant (the maximum frequency). The feedback circuit  25  is the same as that shown in  FIG. 10 . 
     The larger the feedback signal VFB, the heavier the load is judged to be, so that increasing the output current such that the output voltage reaches the target voltage setting is difficult, and so the switching frequency is raised so as to enable accommodation of large changes in the load current. And the smaller the feedback signal VFB, the lighter the load with a small output current is judged to be, so that the switching frequency is set low. 
     When the feedback signal VFB is smaller than a prescribed value (for example 0.4 V), switching is stopped, and a feedback signal VFB voltage higher than the above prescribed value of 0.4 V is awaited. No switching is performed, so that electric charge is not supplied to the secondary-side output capacitor C 3 , and current is supplied only to the load, so that the output voltage falls. As a result the difference between the output voltage and the voltage setting increases, and the voltage value of the feedback signal VFB rises. 
       FIG. 14  shows the configuration of the signal inversion circuit  23  of the switching power supply device shown in  FIG. 13 . The signal inversion circuit  23  comprises an operation amplifier circuit  26 , resistors R 11  and R 12 , and a reference voltage supply E 5 , as shown in  FIG. 14 . 
     Here, the feedback signal VFB is supplied from the feedback circuit  25  via the input terminal FB as a voltage signal of 1 to 2 V, suitable for the positive detection method. The signal is inverted and amplified by the signal inversion circuit  23 , to be converted into an internal signal VFB 2  of 2 to 1.5 V conforming to the negative detection method. The voltage values used by the signal inversion circuit  23  are examples used to explain the range of values which signals may take, and signals are not limited to these values. 
       FIG. 15  shows the configuration of the level shift circuit  24  in the switching power supply device shown in  FIG. 13 . The level shift circuit  24  comprises a resistor R 13  for protection from static electricity and a series circuit of resistors R 14 , R 15  for voltage division, connected between the internal reference voltage E 6  and the signal input terminal for current detection IS, as well as Zener diodes D 5 , D 6  which ground the connection point between the resistors R 13  and R 14 . Here, the negative-voltage current detection signal (the signal itself is a voltage) VIS applied to the signal input terminal IS outputs to the current comparator  17  as an internal signal VIS 2 , which has been level-shifted to a positive potential, from the connection point of the resistors R 14  and R 15 . 
     In this way, the current detection signal VIS is supplied to the signal input terminal for current detection IS as a negative voltage (0 to −1 V); because the IC circuit, which does not have a negative-voltage supply, cannot actually handle a negative-voltage signal, the level shift circuit  24  of  FIG. 15  shifts the signal level to a positive potential (2 to 1.5 V). 
     At this time, the resistance values of the resistors R 11 , R 12  and the like are adjusted such that the output level conforms to this current detection signal, even for the signal inversion circuit  23  which processes the feedback signal VFB. 
     Next, the function of the correction resistor Rc is explained. The correction resistor Rc adds a positive (positive voltage) offset voltage (bias) to the current detection signal VIS, so that in effect the switching frequency determined by the integrated circuit IC is lowered, in order to reduce the power consumption during light loading or no loading or the standby power during standby. Below, the manner in which the correction resistor Rc lowers the switching frequency is explained. 
       FIGS. 16(A)-16(C)  are signal waveform diagrams explaining the correction operation of the current detection signal VIS in a switching power supply device. Here, the signal VFB 3  is a hypothetical signal used for explanation, and is equivalent to the above-described internal signal VFB 2 , with operating range in the positive voltage range (for example 2 to 1.5 V), level-shifted such that the upper limit is 0 V to conform with the operation range of the current detection signal VIS, which is a negative voltage. It may be regarded as the result of inversion of the feedback signal VFB. 
     Here it is assumed that the oscillation frequency of the voltage controlled oscillator  14  is controlled by the feedback signal VFB supplied to the power supply control circuit  10 . 
     First, as shown in  FIG. 16(A) , cases in which correction by the correction resistor Rc is not performed are considered. At this time, the turn-on time ratio of the switching element Q 1  and the value of the feedback signal VFB are in a state of balance such that the voltage output to the load Vout is at the voltage setting. The switching frequency is then determined by the magnitude of the feedback signal VFB. 
     Next, suppose that the correction resistor Rc is added and correction is suddenly applied to the state in  FIG. 16(A) . In this case, the current detection signal VIS is a signal which starts to decline from a larger positive voltage than that in  FIG. 16(A) . On the other hand, the feedback signal VFB, that is, the signal VFB 3  in the figure, cannot change rapidly, so that the same voltage level continues for a time. The switching element Q 1  is not turned off until the current detection signal VIS reaches VFB 3 , so that as shown in  FIG. 16(B) , the turn-on time ton of the switching element Q 1  is lengthened (the turn-on time ton is the interval from the time at which the current detection signal VIS begins to decline until the signal VFB 3  is reached). At this time, if the switching frequency remains unchanged, then the turn-off time within one period is shortened, and the turn-on ratio of the switching element Q 1  is increased. As a result, the voltage output to the load rises, the feedback signal VFB is reduced, and the absolute value of the feedback signal VFB 3  is also reduced. 
     When the feedback signal VFB becomes small, the switching frequency declines, and the time ratio falls, so that the initial turn-on time ratio shown in  FIG. 16(A)  is approached. Hence as shown in  FIG. 16(C) , there is balancing at a new switching frequency, and finally it becomes the same turn-on time ratio as in  FIG. 16(A) . At this time, the feedback signal VFB and the absolute value of the shifted voltage value VFB 3  are smaller than the values before correction. In this way, the frequency controlled by the voltage controlled oscillator  14  goes lower, and the turn-on time determined by the current comparator  17  is also lengthened. 
     In the above-described negative-detection type switching power supply device, current flowing in the correction resistor Rc during light loading remains a problem with respect to promoting energy efficiency. That is, because the correction resistor Rc in a switching power supply device of the prior art is connected to the power supply terminal VCC, current always flows from the power supply terminal VCC through the correction resistor Rc and the series circuit of the resistors Ra and Rs, and through the correction resistor Rc and the series circuit of the resistor Rb and auxiliary windings Lb, to ground (GND), so that there is the problem of the occurrence of power losses. 
     Offsets from two sources are applied to the current detection signal VIS: one from the output voltage of the auxiliary windings Lb via the resistor Rb, and another from the voltage of the power supply terminal VCC via the correction resistor Rc. The voltage of the power supply terminal VCC is proportional to the output voltage Vout, and the output voltage Vout is controlled so as to be a constant voltage, so that the voltage of the power supply terminal VCC is also a constant voltage. On the other hand, because the output voltage of the auxiliary windings Lb is proportional to the input voltage V IN , the value essentially fluctuates. Hence there is the problem that the correction resistor Rc has a complex effect on overcurrent detection. 
     That is, overcurrent detection is performed by comparing the voltage signal from the current detection input terminal IS to a certain reference voltage; however, it is difficult to adjust the circuit constants such that the comparison provides a constant result, regardless of the value of the input voltage V IN . This is because the voltage of the auxiliary windings Lb, which is proportional to the input voltage V IN , is applied to one end of the resistor Rb, and this resistor Rb is connected to the input terminal IS in parallel with the correction resistor Rc, to one end of which is applied a constant voltage (the power supply voltage VCC of the power supply control circuit  10 , a regulated voltage), so that both affect the current detection signal VIS, and constantly adjustment of circuit constants is difficult. 
     This invention has been made in light of the above problems, and has as an object of the provision of a switching power supply device in which, when an external correction circuit is added and the switching frequency during light loading is adjusted from outside, losses in the correction circuit are reduced as compared with the prior art, and moreover adjustment is possible without affecting overcurrent limits or other characteristics. 
     Further objects and advantages of the invention will be apparent from the following description of the invention. 
     SUMMARY OF THE INVENTION 
     In order to resolve the above problems, this invention provides a switching power supply device, comprising: a DC power supply; a transformer which supplies power to a secondary-side load according to a voltage setting; a switching element which is connected in series to primary windings of the transformer; a feedback circuit which outputs to the primary side of the transformer based on a feedback signal obtained by amplifying the difference between the voltage setting and the voltage output to the load; switching power supply control means for performing on/off control of the switching element based on the feedback signal; and current detection means for detecting the value of current flowing in the switching element as a voltage signal with polarity with which as the current value increase, the voltage relative to the reference potential of the switching power supply control means is reduced, the DC power supply being connected to the primary windings of the transformer, the DC voltage input from the DC power supply being turned on and off by the switching element to generate a pulsating current, and the desired output power being supplied to the load according to the voltage setting. 
     The switching power supply control means comprises a signal input terminal for current detection to which the voltage signal is input; a current comparator which compares the feedback signal from the feedback circuit with the voltage signal; an oscillation circuit which, when loading is judged to be light based on the magnitude of the feedback signal, lowers the operating frequency such that the switching interval of the switching element is lengthened; a control circuit which generates a control signal having a frequency and pulse width according to the operating frequency of the oscillation circuit and the comparison result of the current comparator, to drive the switching element; and an output terminal which outputs the control signal, the switching power supply device further comprising a correction circuit, which acts only during on intervals of the switching element, and which applies a positive offset voltage to the voltage signal output from the current detection means. 
     In this switching power supply device, correction is performed by adding an offset to the voltage signal used to detect current values by the correction circuit only during intervals in which the switching signal turns on the switching element. 
     By means of this invention, losses occurring in the correction circuit occur only during control signal on intervals, so that power losses can be reduced compared with methods of the prior art. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram showing an example of a switching power supply device of the invention. 
         FIG. 2  is a block diagram showing the switching power supply device of the first aspect, comprising a correction circuit employing a resistor. 
         FIGS. 3(A) and 3(B)  show operating waveforms of current comparator input signals in the switching power supply device of  FIG. 2 . 
         FIG. 4  is a block diagram showing the switching power supply device of the second aspect, comprising a correction circuit employing a resistor and a capacitor. 
         FIGS. 5(A) and 5(B)  show the operating waveforms of current comparator input signals in the switching power supply device of  FIG. 4 . 
         FIG. 6  is a block diagram showing the switching power supply device of the third aspect, comprising a correction circuit employing a capacitor. 
         FIG. 7  is a block diagram showing the switching power supply device of the fourth aspect, comprising a correction circuit employing a resistor, a capacitor, and a Zener diode. 
         FIGS. 8(A)-8(D)  show operating waveforms of different portions of the switching power supply device of  FIG. 7 . 
         FIG. 9  is a block diagram showing the control circuit of a quasi-resonant switching power supply disclosed in Japanese Patent Laid-open No. 2007-215316 (page 1, line 15 to page 13, line 16, FIG. 4, FIG. 5). 
         FIG. 10  is a block diagram showing one example of a positive-detection type switching power supply device of the prior art. 
         FIGS. 11(A) and 11(B)  show primary-side current waveforms corresponding to different input voltages. 
         FIG. 12  shows changes in the inductor current during overcurrent limiting operation in the switching power supply device of  FIG. 10 . 
         FIG. 13  is a block diagram showing an example of a negative-detection type switching power supply device of the prior art. 
         FIG. 14  shows the configuration of the signal inversion circuit in the switching power supply device of  FIG. 13 . 
         FIG. 15  shows the configuration of the level shift circuit in the switching power supply device of  FIG. 13 . 
         FIGS. 16(A)-16(C)  are signal waveform diagrams explaining an operation to correct the current detection signal VIS in a switching power supply device. 
     
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
     Hereinafter, the invention is explained referring to the drawings. 
     Embodiment 1 
       FIG. 1  is a block diagram showing an example of a switching power supply device of the invention, and  FIG. 2  is a block diagram showing the switching power supply device of Embodiment 1, comprising a correction circuit employing a resistor. 
     First, the switching power supply device comprising a correction circuit  2  employing only a resistor R 0 , as shown in  FIG. 2 , is explained. Here, the correction circuit  2  is connected as an external circuit to the integrated circuit IC. 
     As shown in  FIG. 2 , the integrated circuit IC comprising the power supply control circuit (power supply control means) has a current comparator  17 , which compares a feedback signal VFB from a feedback circuit  25  with a voltage signal from a current detection input terminal IS; a voltage controlled oscillator  14 , which, when the load is judged to be a light load based on the magnitude of the feedback signal VFB, lowers the operating frequency such that the switching intervals of the switching element Q 1  are lengthened; and, a flip-flop circuit  19 , which generates control signals having a frequency and pulse width corresponding to the operating frequency of the voltage controlled oscillator  14  and the comparison result of the current comparator  17 , to drive the switching element Q 1 . The power supply control circuit  10  also has a signal inversion circuit  23  and a level shift circuit  24 . These are the same as those shown in  FIG. 13 , and so a detailed explanation is omitted. The resistor R 0 , serving as the correction circuit  2 , is connected between the output terminal OUT of the power supply control circuit  10  and the signal input terminal for current detection IS. 
     Similarly to that shown in  FIG. 13 , the switching power supply device of  FIG. 1  is a negative-detection type switching power supply device. On the other hand, the switching power supply device shown in  FIG. 13  differs in that a correction circuit  1  is connected such that a positive offset voltage is applied to the current detection signal VIS output from the sense resistor Rs, which is the current detection means, and is configured so as to act only during on intervals of the switching element Q 1 . Otherwise the configuration is as explained for  FIG. 13 ; hereinafter, corresponding portions are assigned the same reference numbers, and explanations are omitted. As the power supply control circuit  10  itself in  FIG. 1 , an integrated circuit IC configured as in the prior art is used. 
     In the case of the switching power supply device of Embodiment 1 shown in  FIG. 2 , current flows in the correction resistor R 0  only during on intervals of the switching element Q 1 , that is, only during intervals in which the switching signal output from the output terminal OUT is high, and no current flows during off intervals. (Strictly speaking, because the potential at the connection point between the resistors Rs and R 0  during off intervals is the result of division of the output voltage from the auxiliary windings Lb by the resistors Rb, Ra, Rs, a small current flows, although it is smaller than the current during on intervals.) Consequently, losses can be reduced compared with the prior art. In particular, the lighter the loading, the smaller is the turn-on time ratio, so that power losses due to current flowing in the correction resistor R 0  can be further decreased. 
       FIGS. 3(A) and 3(B)  show operating waveforms of input signals to the current comparator  17  in the switching power supply device of  FIG. 2 . 
     In  FIG. 3(A) , a current detection signal (voltage signal) during heavy loading is shown. This current detection signal is a voltage signal of polarity which declines with respect to a reference potential (in this case, ground potential) of the switching power supply device as the current flowing in the switching element Q 1  increases, and upon reaching the judgment reference voltage VFB 3  of the current comparator  17  (the same as the hypothetical signal VFB 3  in the explanation of FIGS.  16 (A)- 16 (C)), and causes the switching element Q 1  to be turned off. During the instant in which a switching signal to turn on the switching element Q 1  has been output but the switching element Q 1  has not yet turned on, correction of the current detection signal by the resistor R 0  begins and the current detection signal takes on a positive value, but when the switching element turns on and a current detection signal with large absolute value corresponding to the heavy load appears, the corrected current detection signal VIS becomes a negative signal in the initial stage. 
     In  FIG. 3(B) , a current detection signal during light loading is shown. Similarly to  FIG. 3(A) , at the instant the switching element Q 1  is turned on, correction of the current detection signal by the resistor R 0  starts, and the current detection signal VIS becomes positive. Because the absolute value of the current detection signal is not large for a light load, the interval during the time wherein the corrected current detection signal VIS is positive is a comparatively longer fraction of the cycle. 
     Here, not only is the absolute value of the judgment reference voltage VFB 3  small, but the switching interval T 2  during light loading is long compared with the switching interval T 1  for heavy loading. 
     During intervals in which the switching element Q 1  is off, there is no correction by the resistor R 0 , so that the current detection signal VIS is at ground potential (0 V). The resistance value of the resistor Rb is higher than the values of the other resistors R 0 , Ra, Rs, and the effect on current values and power consumption are smaller than for other resistors, and so to simplify the explanation, the effect of the resistor Rb is ignored (similarly in  FIG. 4  and  FIG. 6  below). The correction circuit  1  need not be an external circuit, but may be incorporated into the integrated circuit IC. 
     Embodiment 2 
       FIG. 4  is a block diagram showing the switching power supply device of embodiment 2, comprising a correction circuit employing a resistor and a capacitor;  FIGS. 5(A) and 5(B)  show operating waveforms of input signals to the current comparator  17  of the switching power supply device of  FIG. 4 . 
     Here, the correction circuit  3  comprises the series circuit of the resistor R 0  and capacitor C 0 . In this case, the time constant determined by the resistor R 0  and capacitor C 0  is made shorter than the switching period T 1  set in the power supply control circuit  10 . 
     In  FIG. 5(A) , the current detection signal (voltage signal) during heavy loading is shown; in  FIG. 5(B) , the current detection signal during light loading is shown. When the switching element Q 1  is turned on, the switching signal changes from low to high, current flows through the capacitor C 0  and resistor R 0  of the correction circuit  3 , and an offset occurs in the current detection signal. Thereafter, charging of the capacitor C 0  ends, and current no longer flows in the correction circuit  3 . In this way, because of the fact that only a current to charge the capacitor C 0  flows in the correction circuit  3  (or from another perspective, because the correction circuit  3  forms a differentiating circuit or a high-pass filter), power losses in the correction circuit  3  occurring during on intervals of the control signal can be further reduced. 
     Overcurrent detection is performed by comparing the current detection signal VIS corrected by the correction circuit  3  with the reference voltage serving as the overload protection (OLP) judgment reference (judgment reference voltage Vth); however, because during the correction interval Tc of the correction circuit  3  the correction circuit  3  operates as a differentiating circuit immediately after turn-on of the switching element Q 1 , there is no effect on the overcurrent limiting action. 
     Embodiment 3 
       FIG. 6  is a block diagram showing the switching power supply device of Embodiment 3, in which the correction circuit in  FIG. 1  employs a capacitor. 
     When the correction circuit  4  comprises only a capacitor C 0 , the operation is substantially the same as the correction circuit  3 , in which a resistor R 0  and capacitor C 0  are connected in series. This is because in the actual correction circuit  4 , parasitic resistance components in the circuit, as well as the sense resistance Rs, or the resistors Ra and Rb connected in series, and other resistance components are equivalent in operation to a resistance component. 
     In the switching power supply devices of Embodiment 2 and Embodiment 3 explained above, the on time is lengthened during heavy loading in which overcurrents may be a problem, so that by means of a configuration comprising a capacitor C 0  as in the correction circuits  3  and  4  shown in  FIG. 4  and  FIG. 6 , by setting the time constant to be shorter than the on time, the effect of the correction circuit  3  or  4  can be eliminated before the current detection signal reaches the judgment reference voltage Vth. As a result, there is no need to consider the effect of the correction circuit  3  or  4  relative to the overcurrent detection level, and only the resistors Rs, Ra, Rb need be considered, so that circuit constant adjustment is simplified. However, the case of a correction circuit  2  comprising only a resistor R 0  ( FIG. 2 ), such as in Embodiment 1, is separate. 
     Embodiment 4 
       FIG. 7  is a block diagram showing the switching power supply device of Embodiment 4, comprising a correction circuit employing a resistor, a capacitor, and a Zener diode. 
     Here, the correction circuit  5  is formed from the resistor R 0 , capacitor C 0 , and Zener diode ZD 0 . In this case, it is desirable that the time constant determined by the resistor R 0  and capacitor C 0  be set to approximately the same as the switching interval T 1  set in the power supply control circuit  10 , or to a length not more than T 1 ; but the time constant may be longer than the switching interval T 1 . The Zener voltage VZD of the Zener diode ZD 0  is set lower than the high level of the driving signal Q 1  output from the output terminal OUT. 
       FIGS. 8(A)-8(D)  show operating waveforms of different portions of the switching power supply device of  FIG. 7 . 
     As shown in  FIG. 8(A) , when the switching element Q 1  is turned on, the voltage of the switching signal from the output terminal OUT of the power supply control circuit  10  changes from low to high (time t 1 ). Then, similarly to the case of the above-described correction circuit  3  (see  FIG. 4 ), a current IC 0  flows through resistor R 0  to the capacitor C 0  of the correction circuit  5 , and this current IC 0  is injected from midway in the series circuit comprising the resistors Rs, Ra, Rb, so that an offset occurs in the current detection signal VIS. Then, the capacitor C 0  is charged, and the voltage at point A in  FIG. 7  rises (see  FIG. 8(B) ). 
     Thereafter, when the voltage at point A reaches the Zener voltage VZD of the Zener diode ZD 0 , the current flowing in the resistor R 0  is shifted from the capacitor C 0  to the Zener diode ZD 0 , and as shown in  FIG. 8(C) , the current no longer flows into the capacitor C 0 . As a result, as shown in  FIG. 8(D) , after time t 2  at which the current flowing in resistor R 0  shifts to the Zener diode ZD 0 , the offset which had been present in the current detection signal VIS due to the correction circuit  5  no longer occurs. 
     Thus in Embodiment 4, by adjusting the Zener voltage VZD of the Zener diode ZD 0 , the time at which an offset occurs in the current detection signal VIS due to the correction-circuit  5  can be freely adjusted. Hence even when the time constant determined by the resistor R 0  and capacitor C 0  is set to be longer than the switching interval T 1 , by selecting the Zener voltage VZD appropriately, the interval in which an offset occurs due to the correction circuit  5  can be set to immediately after turn-on of the switching element Q 1 , and as a result there is no effect on the overcurrent limiting action. Hence compared with the correction circuit  3  of  FIG. 4 , the resistor R 0  and capacitor C 0  can be selected from a broader range of resistance values or capacitance values, and adjustment is made easier. 
     The disclosures of Japanese Patent Applications No. 2007-307743 filed on Nov. 28, 2007 and No. 2008-114705 filed on Apr. 25, 2008 are incorporated as a reference. 
     While the invention has been explained with reference to the specific embodiments of the invention, the explanation is illustrative and the invention is limited only by the appended claims.