Abstract:
Methods and apparatus are provided for stabilizing the phase and gain of an amplifier in a power pooling arrangement. A pilot signal combining a reference carrier frequency with a reference sideband frequency is passed through the amplifier via an adjustable phase shifter and an adjustable attenuator in the amplifier input line. A portion of the amplified pilot signal is coupled from the output of the amplifier and mixed with the reference frequency. The remaining audio frequency signal is compared to a reference (input) audio signal to determine phase and gain changes during amplification in the amplifier. Phase and gain control signals are developed from this comparison, and are fed back to the adjustable phase shifter and adjustable attenuator, respectively, in order to compensate an incoming signal for phase and gain variations in the amplifier.

Description:
TECHNICAL FIELD 
   The present invention generally relates to satellite-based communications systems, and more particularly relates to the stabilization of the gain and phase characteristics of amplifiers used for satellite transmissions. 
   BACKGROUND 
   Satellite-based communication systems are typically used to provide TV broadcast transmissions as well as other types of communication signals over a large geographical area, such as the contiguous 48 states of the U.S. In a typical satellite-based communication system, the satellite receives signals from a feeder Earth station and then re-transmits the signals to end user stations throughout a target area. Ideally, the satellite transmissions are intended to deliver an equitable level of signal power, known as the effective isotropic radiated power (EIRP), to every end user station within the satellite coverage area. One obstacle in achieving ideal EIRP coverage, however, is the adverse effect of localized signal path losses that are typically caused by weather-related conditions such as rain. That is, rain can attenuate transmitted signal strength to a degree that may result in poor quality reception or even a total loss of signal to an end user. 
   To overcome this problem, some satellite transmission systems have been equipped with power-distribution techniques that can enable the individual adjustment of power to the satellite output feed horns in order to boost the amount of power being transmitted to rain-affected areas. One power-distribution technique utilizes signal amplifiers connected to hybrid matrices in order to share amplifier resources for a number of feed horns. In this type of power pooling arrangement, the amplifiers are generally configured to have identical gain and phase parameters in order to achieve optimal performance of the system. However, if these gain and phase parameters are not held constant, cross talk can occur between the output ports, and a loss of output power may result. 
   The stability of the amplifier gain and phase parameters can be adversely affected by influences such as changes in temperature, aging of associated internal and external components, power supply voltage drift, and so forth. In addition, the phase and gain parameters of the line lengths between the amplifiers and the hybrid matrices can also affect the performance of the power pooling arrangement. 
   Accordingly, it is desirable to provide a phase and gain stabilization system that maintains constant phase and gain parameters for amplifiers in a power pooling arrangement over the frequency band of interest. In addition, it is desirable to provide an adjustment capability to compensate for phase and gain differences that may be caused by unequal line lengths between the amplifiers and their associated components. Furthermore, other desirable features and characteristics of the present invention will become apparent from the subsequent detailed description and the appended claims, taken in conjunction with the accompanying drawings and the foregoing technical field and background. 
   BRIEF SUMMARY 
   According to various exemplary embodiments, devices and methods are provided for stabilizing the phase and gain parameters of an amplifier, such as the type for use in a satellite transmitter power pooling arrangement. One method utilizes a pilot signal comprising a reference carrier signal and a reference sideband signal within the operating bandwidth of the amplifier. The pilot signal is introduced into the amplifier and a sample of the amplified pilot signal at the amplifier output is mixed with a reference carrier to extract the sideband signal. The sideband signal is divided into two signals, which are converted into a phase control signal and a gain control signal. The phase control signal and the gain control signal are fed back to the amplifier input circuit to modify the phase and amplitude, respectively, of an input signal to the amplifier, in order to compensate for phase and gain parameter changes within the amplifier. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention will hereinafter be described in conjunction with the following drawing figures, wherein like numerals denote like elements, and 
       FIG. 1  is an exemplary block diagram of a satellite transmitter subsystem; 
       FIG. 2  is an exemplary graph of port-to-port isolation values versus phase error magnitude; 
       FIG. 3  is an exemplary graph of port-to-port isolation values versus amplitude error magnitude; and 
       FIG. 4  is an exemplary embodiment of a phase and gain stabilization network for a TWT amplifier. 
   

   DETAILED DESCRIPTION 
   The following detailed description is merely exemplary in nature and is not intended to limit the invention or the application and uses of the invention. Furthermore, there is no intention to be bound by any expressed or implied theory presented in the preceding technical field, background, brief summary or the following detailed description. 
   Various embodiments of the present invention pertain to the area of satellite communications where a satellite transmitter sub-system utilizes a multi-amplifier power pooling arrangement for individual power adjustment to the respective feed horns. Unless all the amplifiers in such an arrangement have essentially identical phase and gain characteristics, cross talk and power losses can occur at the output ports, thus reducing the efficiency of the transmitter sub-system. To overcome this problem, a phase and gain stabilization network is disclosed herein that operates to maintain essentially constant and identical phase and gain characteristics for each amplifier in a resource-sharing configuration. 
   A typical power pooling transmitter sub-system  100  for a communication satellite (not shown) is illustrated in simplified block diagram form in  FIG. 1 . An input communication signal  102  is typically distributed through a bank of commandable attenuators  104  to a beam-forming network  106 . In general, beam-forming network  106  is made up of power dividers and phase shifters that divide the communication signal into a desired number of inputs for an input hybrid matrix  108 . The corresponding outputs of hybrid matrix  108  are connected to the input ports of a group of amplifiers  110 , which are shown in a power pooling configuration. As previously noted in the Background, power pooling arrangements can be used to enable individual feed horn power control to compensate for particular weather-related signal path attenuations, as described in U.S. Pat. No. 6,421,528. The amplified signal outputs of amplifiers  110  are typically connected to an output hybrid matrix  112  that distributes the communication signal components to a feed horn assembly  114 . A description of this type of hybrid matrix-to-amplifier-to-hybrid matrix arrangement can be found in U.S. Pat. No. 4,618,831. The outputs of feed horn assembly  114  are typically directed to an antenna  116  for broadcast to target areas within the satellite coverage area. 
   In general, the type of amplifiers appropriate for use in satellite transmitter sub-systems are characterized by high gain and wide bandwidth capabilities. One type of amplifier possessing these characteristics is the traveling wave tube, or TWT. The TWT has long been used in many types of RF and microwave applications, such as radar and telecommunications, and is currently used as a signal amplifier in satellite communication applications. Basically, the TWT achieves amplification of an RF or microwave signal through the interaction of a cathode-generated electron beam and a slow wave structure (e.g., a helix) within the TWT. For example, an input RF signal such as a TV broadcast signal that is applied to a TWT helix will typically interact with the energy of the cathode-generated electron beam in a manner that results in amplification of the RF signal at the TWT output. 
   As previously noted, the TWT is typically characterized as a high gain, wide bandwidth device, which is generally advantageous for communications applications. However, the phase shift in a TWT is typically sensitive to changes in the cathode-to-helix voltage, which determines the electron beam velocity. For example, a fractional phase change is typically equal to the negative one-half of a fractional change in the beam voltage. In the case of a TWT having a nominal 4,800 degrees of phase shift, a one-quarter of one per cent change in beam voltage would typically result in six (6) degrees of phase change. In a satellite transmitter sub-system application, six (6) degrees of phase change can be considered a reasonable upper limit in regard to cross talk. However, a typical TWT power pooling configuration may include individual power supplies for each TWT, and the relative voltage drift between the various power supplies may lead to an excessive amount of phase variation. 
   In addition to the previously described effect of beam voltage on phase shift changes, there is also an effect on the gain characteristic of a TWT due to aging of the cathode. As a cathode ages, the electron beam current typically varies, affecting the gain of the TWT. For example, a gain change of one tenth of a neper, or approximately 0.9 db, would have a similar effect on cross talk as would six (6) degrees of phase change. When phase or gain changes (errors) occur in a TWT power pooling arrangement, the amplified output signals may couple to unintended output ports, thereby leading to cross talk and output power losses. 
   Typical examples of port-to-port isolation values versus phase and gain error magnitude are shown graphically in  FIGS. 2 and 3 , respectively. In  FIG. 2 , port-to-port isolation in db is plotted on the vertical axis, and peak-to-peak phase error in degrees is plotted on the horizontal axis. The relationship between port isolation and phase error can be determined by the expression:
 
 C/I (db)=41.14−20*log[ ph pp  (deg)]
 
Where C/I is the ratio of carrier signal to interference signal, as represented by port-to-port isolation.
 
   Similarly,  FIG. 2  depicts port-to-port isolation plotted against peak-to-peak amplitude error in accordance with the expression:
 
 C/I (db)=24.80−20*log[amp  pp  (db)]
 
   As noted above, the effect on port-to-port isolation of a gain error of 0.9 db would approximately equal the effect of a phase error of six (6) degrees. This relationship can be seen on the plots of  FIGS. 2 and 3 , where point A in  FIG. 2  (6 degrees) and point B in  FIG. 3  (0.9 db) each correspond to a port-to-port isolation value of approximately 26 db. 
   In order to control this type of phase and gain error from exceeding levels that would degrade the operational efficiency of a power pooling system, an active phase and gain stabilization network can be configured for each of the amplifiers in the power pooling system. One such stabilization network uses a low power pilot signal (i.e., relative to the power level of the main communication signal, herein designated as “broadband”) at the approximate center frequency of the operating band of the amplifier. In the exemplary embodiment described below in conjunction with  FIG. 4 , an audio signal (e.g., 1 KHz) is used as the sideband on a relatively high frequency carrier. With this approach, the amplitude and phase comparison with a reference audio signal can be done in a narrow bandwidth after demodulation. The use of a narrow bandwidth minimizes the impact of noise in determining the necessary amplitude and phase corrections. The pilot signal is typically introduced into the amplifier in conjunction with the broadband signal, and the phase and gain of the amplified pilot signal output are detected and converted into phase and gain control signals, respectively. These phase and gain control signals can then be used as compensating feedback signals in order to stabilize the phase and gain characteristics of the amplifier. By setting the pilot signal to the approximate center of the operating frequency band, the errors that are typically associated with variations in gain slope and phase slope between the various amplifiers in a power pooling arrangement tend to be minimized. 
   According to an exemplary embodiment of a phase and gain stabilization network  400 , as shown in block diagram form in  FIG. 4 , a TWT amplifier  402  is incorporated within an active stabilization loop  401 . In addition, an amplitude adjust device  404  (typically a commandable attenuator) and a phase adjust device  406  (typically a commandable phase shifter) are inserted in series with the broadband signal input line to active stabilization loop  401 . As such, amplitude adjust device  404  and phase adjust device  406  are generally configured to receive command inputs that enable separate gain and phase adjustments, respectively, to compensate for unequal line lengths in the connections between TWT  402  and the hybrid matrices (not shown) in a typical power pooling arrangement. Amplitude adjust device  404  will generally have an attenuation range of approximately 0 to 4 db, while phase adjust device  406  will typically have a phase shift range of approximately plus or minus 180 degrees. Amplitude adjust device  404  and phase adjust device  406  may be of analog or digital design. If they are configured as digital devices, the least significant bit for amplitude adjust device  404  will generally be 0.25 db, and for phase adjust device  406 , the least significant bit will generally be 3 degrees. 
   In the exemplary embodiment shown in FIG,  4 , an RF reference generator  408  outputs carrier frequency signals at approximately 19.95 GHz that are centered within the operating frequency band (19.7 to 20.2 GHz) of TWT  402 . In addition, an audio reference generator  410  outputs quadrature (cosine and sine) audio frequency signals at approximately 1 KHz. In this embodiment, a single sideband modulator configuration, as described below, derives a pilot signal from a combination of the carrier frequency signal and the audio frequency signal, and this pilot signal is used for detecting phase and gain variations in TWT  402 . That is, a 19.95 GHz output signal from reference generator  408  is passed through a 90-degree hybrid  412 , which has a load  414  on one port. Hybrid  412  outputs quadrature (cosine and sine) 19.95 GHz reference signals to balanced modulator  415  and balanced modulator  416 , respectively. Concurrently, audio reference generator  410  outputs 1 KHz quadrature (cosine and sine) signals to balanced modulator  416  and balanced modulator  414 , respectively. The signal outputs of balanced modulators  415  and  416  are combined in a sum device  418  to provide a single sideband suppressed carrier signal at 19,950.001 MHz, herein designated as a pilot signal. 
   The exemplary pilot signal described above, typically set at a power level of approximately one milliwatt, is coupled into the broadband input line of TWT  402  via a directional coupler  420 . A typical coupling factor for directional coupler  420  is −30 db, thereby resulting in a one-microwatt power level pilot signal at the output of directional coupler  420 . The one-microwatt pilot signal is then directed through a voltage controlled attenuator  422  and a voltage controlled phase shifter  424  prior to reaching the input of TWT  402 . As will be described below, voltage controlled attenuator  422  and voltage controlled phase shifter  424  are typically used to apply compensating gain and phase adjustments to the broadband signal in the input line to TWT  402 . 
   The exemplary pilot signal is amplified by TWT  402  and passed through a directional coupler  426 . Assuming a nominal amplification of 55 db for TWT  402  and a nominal coupling factor of −55 db for directional coupler  426 , the pilot signal level at the output of directional coupler  426  is approximately one microwatt, which is about 20 db below the level of the coupled broadband signal. The coupled output from directional coupler  426  is fed to a balanced mixer  428 , where it is mixed with a local oscillator signal (19.95 GHz) from RF reference generator  408  at an approximate power level of 3 milliwatts. The output signal from balanced mixer  428  typically contains a 1 KHz signal at a power level of approximately one-quarter microwatt, and also approximately 25 microwatts of the broadband signal. For a balanced mixer  428  nominal output impedance of 50 ohms, the 1 KHz output signal will typically be at a voltage level of approximately 3.5 millivolts. The output signal from balanced mixer  428  is fed to a stabilized audio amplifier  430 , which typically has a gain of 50 db and a bandwidth of 1 KHz, with a center frequency of approximately 1 KHz. The resulting output from audio amplifier  430  typically contains a 1 KHz signal at a voltage level of approximately one volt, plus incidental noise associated with the broadband signal of only about 10 millivolts in the narrow 1 KHz band. 
   The 1 KHz signal out of stabilized audio amplifier  430  is fed into a pair of phase detectors,  432  and  434 . Concurrently, phase detectors  432 ,  434  receive quadrature reference signals at 1 KHz from stabilized audio reference generator  410 . In the exemplary embodiment of  FIG. 4 , phase detector  432  receives a cosine (90 degrees) output from audio reference generator  410 , and phase detector  434  receives a sine (0 degrees) output from audio reference generator  410 . The signal output of phase detector  432  is typically a DC voltage that is proportional to the phase difference between the 1 KHz signal input from stabilized audio amplifier  430  (representing the output pilot signal) and the 1 KHz reference signal from audio reference generator  410  (representing the input pilot signal). This DC voltage typically represents a change (error) in the phase parameter of TWT  402 , and can therefore be used in a feedback loop to correct for phase changes in TWT  402 . The DC voltage out of phase detector  432  is typically fed through an integrator  436  prior to being applied to voltage controlled phase shifter  424 . As such, the signal output of integrator  436  represents a relatively high gain feedback signal for correcting phase changes in TWT  402 . The feedback signal out of integrator  436  is typically applied to voltage controlled phase shifter  424  in a manner that adjusts the phase of the broadband signal passing through voltage controlled phase shifter  424  to compensate for the phase shift error detected in TWT  402 . The parameters of this phase control feedback loop are typically designed to drive the output of phase detector  432  to zero, so that phase changes in TWT  402  are canceled by opposite phase changes in voltage controlled phase shifter  424 . 
   The function of phase detector  434  is similar to that of phase detector  432 , except that the output signal from phase detector  434  is typically proportional to the gain of TWT  402 , and can be used as a gain control feedback signal. In this embodiment, the output signal from phase detector  434  is compared to a predetermined reference voltage in the comparator section of a comparator and integrator  438 , where the predetermined reference voltage represents a desired gain for TWT  402 . The difference voltage between the phase detector  434  output voltage and the predetermined reference voltage is typically integrated by the integrator section of comparator and integrator  438 , and applied as a gain control signal to voltage controlled attenuator  422 . This gain control signal is typically applied to voltage controlled attenuator  422  in a manner that adjusts the amplitude of the broadband signal passing through voltage controlled attenuator  422  to compensate for the gain error in TWT  402 . As such, this gain control feedback loop is typically designed to stabilize the net gain of the TWT  402  circuit to a desired level. 
   Accordingly, the shortcomings of the prior art have been overcome by providing an improved phase and gain stabilization network for an amplifier, such as the type of amplifier that is generally used in a satellite transmitter power pooling arrangement. The exemplary embodiment of a phase and gain stabilization network described herein can maintain essentially constant phase and gain parameters for the signal amplifier circuits in power pooling arrangements, such as those that may be used in satellite applications. It will be appreciated that other embodiments of the phase and gain stabilization network described herein may be used for other types of parallel amplifier summing applications (e.g., terrestrial applications), and may be used for other frequency bands as well. Moreover, the exemplary embodiment also provides commandable adjustment capabilities to compensate for phase and gain differences caused by unequal line lengths between the amplifiers and their associated components. 
   While at least one exemplary embodiment has been presented in the foregoing detailed description, it should be appreciated that a vast number of variations exist. It should also be appreciated that the exemplary embodiment or exemplary embodiments are only examples, and are not intended to limit the scope, applicability, or configuration of the invention in any way. Rather, the foregoing detailed description will provide those skilled in the art with a convenient road map for implementing the exemplary embodiment or exemplary embodiments. It should be understood that various changes can be made in the function and arrangement of elements without departing from the scope of the invention as set forth in the appended claims and the legal equivalents thereof.