Abstract:
Methods and apparatus for amplifying a tuner input signal are disclosed. One embodiment of the invention is directed to a tuner amplifier system comprising a tuner amplifier input that receives a tuner amplifier input signal and a first amplifier comprising an input and an output. The input of the first amplifier is coupled to the tuner amplifier input. The system further comprises a second amplifier comprising an input and an output, the input of the second amplifier being coupled to the tuner amplifier input, and a switch adapted to couple one of the first amplifier output and the second amplifier output to an output of the tuner amplifier. Another embodiment of the invention is directed to a method of amplifying a tuner input signal. The method comprises acts of detecting a power of the tuner input signal, selecting a tuner amplifier to amplify the tuner input signal based on the power of the tuner input signal, and amplifying the tuner input signal using the selected amplifier.

Description:
FIELD OF THE INVENTION 
   The present invention is directed generally to the field of tuners, and more particularly to tuners implemented in silicon. 
   DESCRIPTION OF THE RELATED ART 
   Television signals are transmitted in many different forms. For example, different regions of the world use different transmission standards, which dictate the parameters of transmitted television signals. In addition, television signals may be transmitted through cables that connect to a television or associated set-top box, or alternatively may be transmitted through the air and received using an antenna or satellite dish. In each case, the transmitted signals may be encoded in analog and/or digital format. The encoded signals may then be modulated into a channel using digital or analog modulation, although some forms of information (e.g., data) may only be modulated using one form of modulation (e.g., digital). In both cases, a bandwidth of 6–8 MHz defines a channel. During transmission of the signal, the channel is placed in a frequency range between 43 MHz and 1 GHz. 
   Each of the television signals described above has different attributes. For example, the width of the channel may vary depending on its transmission standard and whether the signal is modulated using digital or analog modulation, and the power of the television signals may differ depending on whether the signal is transmitted using a cable or terrestrial broadcast and whether the signal is modulated using digital or analog modulation. Thus, each type of television signal dictates a different set of tuner specifications. As a result, a television or set-top box must be provided with a separate tuner for each different type of television signal it is equipped to receive. Providing multiple tuners in a television or set-top box is space and cost intensive. 
   In view of the foregoing, one object of the invention is directed to a tuner adapted to receive and process different types of television signals, such as analog modulated cable signals, digitally modulated cable signals, analog modulated over-the-air (or “terrestrial”) signals, digitally modulated terrestrial signals and/or signals having different transmission standards. 
   SUMMARY OF THE INVENTION 
   One embodiment of the invention is directed to a tuner amplifier system. The system comprises a tuner amplifier input that receives a tuner amplifier input signal and a first amplifier comprising an input and an output. The input of the first amplifier is coupled to the tuner amplifier input. The system further comprises a second amplifier comprising an input and an output, the input of the second amplifier being coupled to the tuner amplifier input, and a switch adapted to couple one of the first amplifier output and the second amplifier output to an output of the tuner amplifier. 
   Another embodiment of the invention is directed to a method of amplifying a tuner input signal. The method comprises acts of detecting a power of the tuner input signal, selecting a tuner amplifier to amplify the tuner input signal based on the power of the tuner input signal, and amplifying the tuner input signal using the selected amplifier. 
   A further embodiment of the invention is directed to a tuner amplifier system comprising a tuner amplifier input that receives a tuner amplifier input signal and a tuner amplifier output that transmits a tuner amplifier output signal. The system further comprises a first amplifier comprising an input and an output, the input of the first amplifier being coupled to the amplifier input, and a second amplifier comprising an input and an output, the input of the second amplifier being coupled to the amplifier input. The tuner amplifier output signal comprises one of a signal from the first amplifier and a signal from the second amplifier at a given time. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIGS. 1A and 1B  are block diagrams of conventional television apparatus; 
       FIG. 2  is a schematic representation of a tuner in accordance with one embodiment of the invention; 
       FIG. 3  is a schematic representation of an exemplary implementation of the first die shown in  FIG. 2 ; 
       FIG. 4  is a schematic representation of an exemplary implementation of the second die shown in  FIG. 2 ; 
       FIG. 5  is a schematic representation of an exemplary implementation of the filter  53  is shown in  FIG. 4 ; 
       FIG. 6  is a schematic representation of an exemplary implementation of the LNA, IMF, and AGC shown in  FIG. 4 ; 
       FIG. 7  is a circuit diagram of an exemplary implementation of a transconductance amplifier, such as the transconductance amplifiers shown in  FIG. 6 ; 
       FIG. 8  is a circuit diagram of an exemplary implementation of the first amplifier shown in  FIG. 6 ; 
       FIG. 9  is a circuit diagram of an exemplary implementation of the second amplifier shown in  FIG. 6 ; 
       FIG. 10  is a schematic representation of an exemplary implementation of the AGC shown in  FIG. 6 ; 
       FIG. 11  is one example of a function that may be implemented by the hysteresis controller shown in  FIG. 10 ; and 
       FIG. 12  is a circuit diagram of an exemplary implementation of the IMF shown in  FIG. 6 . 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   A television tuner is a device that receives a television signal from a cable or terrestrial source and outputs a channel of interest, rejecting all other channels. For example, the television tuner may receive a plurality of channels having a signal frequency between 43 MHz and 1 GHz. The tuner may output a single such channel having a bandwidth between approximately 6 MHz and 8 MHz. 
   Conventional tuners are adapted to receive either digitally modulated or analog modulated signals.  FIGS. 1A and 1B  illustrate examples of a digital tuner and an analog tuner, respectively, in the context of conventional television apparatuses. As shown in  FIG. 1A , a digital channel tuner  1   a  receives digitally modulated television signals from a cable or terrestrial source  3 . The digital channel tuner  1   a  outputs a channel of interest, which is demodulated by digital channel demodulator  2   a  and decoded by an MPEG decoder  5 . Next, a graphics processor  4  processes the graphics of the channel for display. Finally, a display  7 , such as a plasma display, liquid crystal display, digital light projection display or cathode ray tube, displays the decoded channel of interest. In  FIG. 1B , an analog channel tuner  1   b  receives analog modulated television signals from a cable or terrestrial source  3 . The analog channel tuner  1   b  outputs a channel of interest, which is demodulated by analog channel demodulator  2   b  and decoded by a video and audio processor  6 . Next, a graphics processor  4  processes the graphics of the channel for display. Finally, as in  FIG. 1A , a display  7  displays the decoded channel of interest. Conventional television apparatus adapted to receive both analog modulated and digitally modulated signals typically require both the digital channel tuner  1   a  of  FIG. 1B  and the analog channel tuner  1   b  of  FIG. 1B . 
   Tuning to a particular channel involves translating signals in frequency using a mixer. As a result, tuners can have the undesired effect of translating both a channel of interest and an unwanted image channel. For example, if a channel of interest is translated to an intermediate frequency IF, an image channel will be also translated to the same intermediate frequency. The image channel interferes with the channel of interest, and must be eliminated to a large degree for proper reception. To eliminate the image channel, tuners may use filters and/or image-reject mixers. The degree to which the image channel is rejected is called “image rejection.” 
   In single conversion tuners, a tracking bandpass filter is used to reject the image channel prior to frequency translation. However, such filters have drawbacks. For example, a tracking bandpass filter requires discrete components with a high quality factor. In addition, because the filter is associated with a high loss, the amplifier following the filter must have a low noise figure. An alternative is a dual conversion architecture. In dual conversion tuners, a signal is upconverted to a first high intermediate frequency (IF), having a typical center frequency in the range of 1 GHz to 2 GHz. The image signal generated during this upconversion is not troublesome as the image signal is outside of the television frequency band. The image signal for the downconversion operation is largely attenuated compared to the signal of interest using filtering at the intermediate frequency. The signal is then downconverted to a second low IF, having a typical center frequency in the range of 36 MHz to 58.75 MHz, and is further filtered to isolate the channel of interest. Filtering at the low IF is typically performed using surface acoustic wave (SAW) filters. 
   Conventional single and dual conversion tuners have a number of limitations. For example, the SAW filters that are used to isolate the channel of interest at the low IF are cumbersome to implement. In addition, these SAW filters are specialized for use with television signals having particular attributes such as a given channel bandwidth. Thus, tuners that use SAW filters to isolate the channel of interest at the low IF typically cannot be used with television signals of different standards and types. As a result, multiple tuners or multiple SAW filters are needed to receive television signals of different standards and types. 
   One aspect of the present invention is directed to a tuner that is adapted to process both analog modulated signals and digital modulated signals. Another aspect of the invention is directed to a tuner that is adapted to process both cable signals and terrestrial signals. A further aspect of the invention is directed to a tuner that is adapted to process signals conforming to different standards. Thus, the tuner may have improved flexibility relative to conventional tuners. Another aspect of the invention is directed to a television tuner that may be implemented in silicon. Thus, the tuner may have reduced cost and/or improved performance. Although these aspects of the present invention may be advantageously employed together, the present invention is not limited in this respect, as each of these aspects of the present invention can also be employed separately or in any combination. 
   According to one embodiment of the invention, the tuner may have a dual conversion architecture, but may perform low IF filtering on digital signals rather than analog signals. In particular, an analog to digital converter may be used to convert the analog signals to digital signals after downcoversion of the signals to the low IF. According to one exemplary implementation, the analog to digital converter may be a sigma delta converter, which provides dynamic range and inherent anti-aliasing, and the mixer that down-converts the signals to the low IF may be a quadrature mixer that generates quadrature signals. Filtering in the digital domain eliminates the need for a SAW filter at the low IF, and therefore may reduce the cost of the tuner. In addition, filtering in the digital domain allows the filtering parameters to be selected based on an aspect of a signal (e.g., the television signal standard), for example by programming the coefficients of a digital filter. 
   According to another embodiment of the invention, the gain of a broadband signal input to a tuner may be controlled based on an aspect such as power of the broadband signal. Thus, the gain of the signal may be controlled based on the power of the entire input signal, including unwanted channels, so that a proper gain may be selected. To control the gain, a control loop having a programmable automatic gain controller may be included around the front end amplifier of the tuner. Controlling the gain of the signal based on an aspect of the signal allows the gain of the signal to be adjusted in accordance with particular tuner applications, such as terrestrial or cable reception. 
   According to a further embodiment of the invention, the tuner may be implemented in a single package or module, such as a multi-chip module, which may include a single substrate. The package or module may include a first die for analog processing and a second die for digital processing. Advantageously, the package or module may be a single unit that has the appearance of a single chip. 
     FIG. 2  illustrates an embodiment of the invention in which a silicon-implemented tuner  9  comprises a first die  11 , which includes circuitry that performs analog processing, and a second die  13 , which includes circuitry that performs digital processing. The silicon-implemented tuner  9  may be implemented as a single package or module that includes the first die  11  and second die  13 . For example, the tuner  9  may be implemented as a multi-chip module in which the first and second dies  11 ,  13  are assembled on a single substrate. 
   The first die receives a television signal  15  from a television signal source  17 , which may be a terrestrial receiver (e.g., a television antenna or satellite dish) or a cable. The signals received from the cable or terrestrial source may be digitally modulated or analog modulated television signals. The analog modulated television signals may be encoded in standard NTSC (National Television System Committee) form, which is the transmission standard used in the United States and Canada. Alternatively, the signals may be encoded using the PAL (Phase Alternating Line) transmission standard used in Western Europe and Australia, the SECAM (Sequential Couleur Avec Memoire or Sequential Colour with Memory) transmission standard used in Eastern Europe and France, or another transmission standard. The digitally modulated television signals may be compressed in accordance with MPEG and Dolby standards and transmitted using VSB (Vestigial Sideband Modulation)/QAM (Quadrature Amplitude Modulation), e.g. in the United States, or OFDM (Orthogonal Frequency Division Multiplexing), e.g., in Japan and Europe. 
   It should be appreciated that although tuner  9  is described herein as receiving and processing television signals, the invention is not limited in this respect. The tuner  9  may be used to receive radio signals or information signals for which frequency tuning is required. Further, the tuner  9  may receive audio, video and/or information signals that are displayed or processed by a device other than a television, such as a personal computer. 
   The television signal  15  is processed by the analog circuitry of the first die  11  and conveyed to the second die  13  as a pair of quadrature signals  19   a–b . The second die  13  digitally processes the quadrature signals  19   a–b  and outputs a channel of interest as a pair of signals  21   a–b , which may be an I/Q quadrature signal. Signals  21   a–b  may then be transmitted to an digital channel demodulator  23  for demodulating the digital channel. In addition, the second die  13  may output signals  22   a–b , which correspond to an audio and video signal, respectively. Signals  22   a–b  may then be transferred to a video and audio processor  24 . 
   The first and second dies  11 ,  13  may use silicon technology suitable for analog and digital processing, respectively. For example, the first die  11 , which includes circuitry that performs analog processing, may use bipolar or bipolar complementary metal oxide semiconductor (BiCMOS) technology. The second die  13 , which includes circuitry that performs digital processing, may use complementary metal oxide semiconductor (CMOS) technology. In addition, the first and second dies  11 ,  13  may be insulated from each other to reduce noise coupling. 
     FIG. 3  illustrates an exemplary implementation of the first die  11  of  FIG. 2 . As shown, the first die  11  includes circuitry that performs gain adjustment, upconversion to a first intermediate frequency, and initial filtering of the television signal  15 . The circuitry then performs downconversion of the filtered signal to a second intermediate frequency and converts the signal to quadrature signals  19   a–b . However, it should be appreciated that it is not necessary that all of these function be performed on the first die  11 . For example, some or all of the filtering of the television signal  15  may be performed external to first die  11 . 
   The gain adjustment of the television signal  15  is performed by low noise amplifier (LNA)  25 , which receives the television signal  15  at an input  27  of the first die  11 . LNA  25  may have a variable gain according to one exemplary implementation. An image filter (IMF)  29  receives the output of LNA  25 , and filters the output signal to reduce the noise contribution of the LNA  25 . The television signal at the output of the LNA  25  may have a decreased or increased amplitude relative to the signal at the input of the LNA  25 . According to one embodiment of the invention, the gain of the LNA  25  may be adjusted based on a factor such as a power of the television signal. For example, the gain of the LNA  25  may be adjusted based on the root mean square (RMS) power or the peak power of a broadband television signal. Thus, the composite power of the entire television frequency band (e.g., 43 MHz to 1 GHz) may be measured, rather than merely the power of the desired channel as in conventional implementations of such gain control loops. In the exemplary implementation of  FIG. 3 , gain control signals are provided by automatic gain controller (AGC)  31  to the LNA  25  based on a detected average power of the full television signal detected at the output of the IMF  29 . Exemplary implementations of LNA  25 , AGC  31 , and IMF  29  will be discussed in detail in connection with  FIGS. 6–11 . 
   The output of the IMF  29  is upconverted to a first intermediate frequency by a mixer  33  coupled to a local oscillator  35 . In one example, the signal is upconverted to an intermediate frequency of approximately 1.9 GHz. The output of mixer  33  is then processed by a filter  37  that eliminates most of the channels other than the channel of interest. For example, the output of the filter  37  may consist of two to four unattenuated channels adjacent to the desired channel, according to one implementation. Local oscillator  35  may be a fractional phase locked loop frequency synthesizer in one example, such as part number ADF4252 manufactured by Analog Devices, One Technology Way, Norwood, Mass. 02062. Filter  37  may be a band-pass filter having a high quality factor so as not to attenuate the channel of interest. According to one implementation, the filter  37  is a SAW filter with a bandwidth of approximately 20 MHz and image rejection of approximately 40 dB. To achieve the desired image rejection, additional image rejection may be performed at another portion of tuner  9 . It should be appreciated that while filter  37  is shown as being included on the first die  11 , filter  37  may alternatively be implemented, at least in part, externally to the first die  11 . Filter  37  is followed by a quadrature mixer  39 , which converts the output of the filter  37  into quadrature signals  19   a–b  having a second intermediate frequency. In particular, mixers  39   a  and  39   b  are coupled to a local oscillator  41  that may be fixed at a frequency lower than that of local oscillator  35  to downconvert the output of the filter  37 . Mixer  39   a  is driven by the in-phase (I) component of the local oscillator  41  signal, while mixer  39   b  is driven by the quadrature component (Q) of the local oscillator  41  signal. Downconverted quadrature signals  19   a–b  are output from the first die  11  via outputs  33   a–b  and may comprise a single-ended or differential quadrature representation of the desired channel. According to one implementation, the dynamic range of the quadrature mixer  39  may be approximately the same as the dynamic range of the mixer  33 , so that a similar input may be accommodated. The local oscillator  41  may be a fractional phase locked loop frequency synthesizer such as part number ADF4252 or an integer phase locked loop frequency synthesizer such as part number ADF4112 manufactured by Analog Devices, One Technology Way, Norwood, Mass. 02062. According to one implementation, the local oscillator  41  has a frequency between approximately 1.8 GHz and approximately 2.0 GHz, although other frequencies are possible. According to one embodiment, the frequency of the local oscillator  41  may be selected or programmed based on the transmission standard of the television signal  15 , as different transmission standards use different intermediate frequencies. Alternatively, the frequency of local oscillator  41  may be fixed. 
   Each of local oscillators  35  and  41  may have a frequency step size greater than that of conventional frequency synthesizers. For example, the step size of local oscillators  35  and  41  may be greater than 62.5 KHz. For example, local oscillator  35  may have a step size of approximately 1 MHz and local oscillator  41  may have a step size of approximately 500 kHz. The resolution requirement for local oscillators  35  and  41  may be relaxed as a result of digital processing performed on second die  13 . Such digital processing may translate the signal of interest to a desired frequency with sufficient resolution so that the step size of local oscillators  35  and  41  may be increased. 
     FIG. 4  illustrates an exemplary implementation of the second die  13  of  FIG. 2 . The die may include circuitry that performs analog-to-digital conversion of quadrature signals  19   a–b , frequency translation of the signals to baseband, and additional filtering. The quadrature signals  19   a–b  output by the outputs  33   a–b  of the first die  11  are input to inputs  45   a–b  of the second die  13 . According to one aspect of the invention, quadrature signals  19   a–b  may be current signals. The current signals may be differential current signals, according to one example, and may be generated by quadrature mixer  39 . The current signals may be transmitted directly to the second die  13  of tuner  9 . In particular, quadrature signals  19   a–b  may be transmitted directly to an analog-to-digital converter (ADC)  47  of the second die  13 . 
   The ADC  47  may convert the analog quadrature signals into digital quadrature signals that are subjected to filtering and decimation. For example, the output of the ADC  47  may be decimated by a factor of eight or sixteen. In addition, some of the channels adjacent to the channel of interest that are not eliminated by filter  37  (FIG.  3 ) may be attenuated by digital filtering following the ADC  47 . In one example, the ADC  47  is a sigma delta converter. A sigma delta converter samples an analog input at many times the Nyquist rate and produces a one or multi-bit output that tracks the analog input in the frequency range of interest. This output is processed by a digital filter to produce a high resolution conversion result. The sigma delta converter may be implemented in a continuous-time fashion, which provides for an anti-aliasing function. Due to the high dynamic range and inherent anti-aliasing function of the ADC  47  implemented as a sigma delta converter, it is not necessary to perform filtering at the second intermediate frequency as in conventional tuners. This results in a tuner having a reduced cost. 
   The output of ADC  47  is translated to baseband, which may be at or near DC (0 Hz), by a quadrature mixer  49   a–b  coupled to numerically controlled oscillators (NCO)  51   a–b . NCOs  51   a–b  enable the output of the ADC  47  to be translated to baseband with sufficient resolution such that the resolution of local oscillators  35  and  41  may be relaxed, as described previously. According to one exemplary implementation, NCOs  51   a–b  have a step size of approximately 62.5 KHz or less. The output of the mixer  49  is then filtered by a filter  53 , which further isolates the channel of interest. One advantage of filtering at baseband is that it is less expensive than filtering at other frequencies. 
   According to one embodiment of the invention, the filter  53  may be programmable to accommodate different types of signals, such as digitally modulated cable signals, digitally modulated terrestrial signals, analog modulated cable signals, or analog modulated terrestrial signals. In addition, the filter  53  may be programmable to accommodate television signals adhering to different standards in use around the world, such as NTSC in the United States and Canada, PAL in Western Europe and Australia, and SECAM in Eastern Europe and France. Thus, the filter  53  may be a Nyquist filter. It should be appreciated that the foregoing examples are merely exemplary, and that the filter  53  may be programmed to accommodate television signals adhering to other standards, such as those recognized by the International Telecommunication Union (ITU). One parameter of the filter  53  that may be modified is the bandwidth of the filter. For example, to program the filter  53  to accommodate a digital cable television signal in the United States, the bandwidth may be set at 6 MHz. To program the filter  53  to accommodate a digital terrestrial television signal in Australia, the bandwidth may be set at 7 MHz. Another parameter of the filter  53  that may be modified is frequency characteristics of the filter. 
   The filter  53  may be programmed with filter coefficients corresponding to a particular signal type and standard, and reprogrammed to accommodate different signal types and standards. According to one example, programming may be performed using a set top box controller or a television controller that transmits control signal  55  to the filter  53 . The control signal  55  may be transmitted to the filter  53  via input  224 , which may for example be coupled to a serial port interface. 
   According to another example, the modulation type of the television signal (e.g., analog or digital) may be detected at filter  53 , and the filter coefficients may be programmed or adapted based on the detected modulation type. Two exemplary ways of detecting the modulation type of a television signal are attempting to detect a carrier signal in the frequency domain, and attempting to detect a sync signal in the time domain. By detecting the presence of an analog signal based on one of these characteristics, it is possible to distinguish analog modulation from digital modulation. Referring to the first example, if a carrier signal is detected in the television signal, it may be concluded that the signal is modulated using analog modulation. If a carrier signal is not detected, it may be concluded that the signal is digitally modulated. Referring to the second example, if a sync signal is detected in the television signal, it may be concluded that the signal is modulated using analog modulation. If a sync signal is not detected, it may be concluded that the signal is digitally modulated. Filter  53  may be adaptive, such that its filter coefficients may be set or changed automatically (e.g., without user intervention) in response to the detection of analog or digital modulation. 
     FIG. 5  illustrates an exemplary implementation of the filter  53  is shown in  FIG. 4 . Signals  208   a–b  from quadrature mixer  49   a–b  are input to the filter  53 . One of signals  208   a  and  208   b  is input to a modulation detector  209  so that modulation detector  209  may detect whether the signal is analog modulated or digitally modulated. Signal  208   a  is shown as input to modulation detector  209  in  FIG. 5 , however signal  208   b  or both signals  208   a  and  208   b  may alternatively be input to modulation detector  209 . Modulation detector  209  may, for example, comprise a carrier detector or a sync detector as described above, or may comprise both a carrier detector and a sync detector. Modulation detector  209  outputs a signal indicative of whether signal  208   a  is analog modulated to each of switches  211  and  233 . This signal controls the state of each of the switches  211  and  233 . In response to the signal received from modulation detector  209  at input  212 , switch  211  selects one of the outputs  217  and  219  to be connected to the input  213  of the switch. If the signal received from modulation detector  209  indicates analog modulation of signals  208   a , the switch  211  outputs the quadrature signals  208   a–b  at output  217 . If the modulation detector  209  does not detect analog modulation of signal  208   a , switch  211  outputs the signals at output  219 . Thus, analog modulated signals are transmitted to carrier demodulator  222  and then to audio/video filter  221 , while digitally modulated signals are transmitted to digital filter  223 . Audio/video filter  221  receives and processes the demodulated signals transmitted from carrier demodulator  222 . Audio/video filter  221  may be programmable via input  224 , such that parameters of the filter may be changed based on a source (e.g., cable or terrestrial) and/or standard (e.g., NTSC, PAL, or SECAM) of television signal. For example, filter coefficients may be received via input  224 . Digital filter  223  may be adapted to process digitally modulated signals and may be programmable via input  224  such that parameters of the filter may be changed based on a source and/or standard of television signal. 
   Inputs  227  and  229  of switch  233  are respectively coupled to audio/video filter  221  and digital filter  223 . In response to the signal received from modulation detector  209  at input  225 , switch  233  selects one of the inputs  227  and  229  to be connected to the output  231  of the switch. In particular, switch  233  may output signals received from audio/video filter  221  if modulation detector  209  detects analog modulation of signal  208   a  and may output signals received from digital filter  223  otherwise. Quadrature signals  230   a–b  output by the switch  233  are transmitted to mixers  57   a–b.    
   Referring again to  FIG. 4 , mixers  57   a–b  and NCOs  59   a–b , coupled thereto, may be optionally included as part of second die  13  to upconvert the output of the filter  53  to a particular frequency (e.g., 44 MHz) so that the tuner output is compatible with a particular standard having such a frequency requirement. The outputs of the mixers  57   a–b  are transmitted to digital tuner outputs  61   a–b  in the form of digital signals  21   a–b , which may be an audio signal and a video signal or a real and imaginary signal. Alternatively, a single real signal or a single signal with both audio and video information may be used. The digital signals  21   a–b  may be provided directly to a digital decoder, such as a digital decoder in a set-top box, without any need for analog-to-digital conversion. The outputs of the mixers  57   a–b  are also provided to digital-to-analog converters (DACs)  63   a–b , respectively, so that analog signals  22   a–b  may be provided at analog tuner outputs  67   a–b . The analog signals  22   a–b , which are analog representations of digital signals  21   a–b , may be provided to a digital decoder having analog inputs. 
   Having described an exemplary implementation of the tuner  9 , the LNA  25 , AGC  31 , and IMF  29  of  FIG. 3  will now be addressed in greater detail. As previously discussed, the LNA  25  may have a variable gain in accordance with one embodiment of the invention. One benefit of having a variable gain is that the gain may compensate for signals of varying power to achieve a substantially constant level at the output of the tuner. Compensating for signals of varying power may be particularly advantageous when the television signals received by the tuner may originate from both cable and terrestrial sources, as cable signals tend to have a much higher power than terrestrial signals. 
   Using a variable gain amplifier, a television signal from a cable source may be attenuated so that further processing by the tuner does not generate distortion in the television signal, and a television signal from a terrestrial source may be amplified so that noise resulting from further processing does not degrade the signal. In addition, using a variable gain amplifier to vary the gain of the television signal based on the composite power of the input signal may reduce the dynamic range requirements of the tuner processing circuitry. Thus, the dynamic range at the output of the LNA  25  may be smaller than the dynamic range at the input of the LNA  25 . In one example, the dynamic range at the input of the LNA  25  is approximately equal to the difference between the maximum signal that can be processed by the tuner with acceptable distortion and the minimum signal that can be processed by the tuner with acceptable noise. At the output of the LNA  25 , the dynamic range may be the difference between the input dynamic range and the gain range of the LNA  25 . 
   An exemplary implementation of the LNA  25 , AGC  31 , and IMF  29  of  FIG. 3  will now be described. According to one embodiment of the invention, LNA  25  is implemented using a plurality of amplifiers each having an architecture that is optimized for the receipt of particular signals. For example, two amplifiers may be used to process cable and terrestrial television signals. For example, one amplifier (e.g., a fixed gain amplifier) may be used to process weaker signals received and another amplifier (e.g., a variable gain amplifier) may be used to process stronger signals received. Weaker signals tend to be terrestrial signals, although this is not always necessarily the case. The impedance of each amplifier may be matched to the impedance of the television signal source (e.g., cable or antenna), and the gain of each amplifier may be selected based on the power of the signals it receives. 
   According to one embodiment of the invention, different impedance matching schemes are used for the receipt of stronger (e.g., cable) and weaker (e.g., terrestrial) signals. Although the impedance of both cable and antenna television signal sources is approximately the same, different considerations apply in selecting the elements for the impedance matching scheme of a cable and antenna. For example, it may be preferred that the impedance matching scheme of the antenna generate very little noise to lessen the degradation of the television signals received by the antenna, as terrestrial signals tend to be relatively weak. It may also be preferred that the impedance matching scheme of the cable be compatible with having a variable gain, as having a variable gain increases the dynamic gain range of the tuner so that signals having a large range of power levels may be received. Noise is less of a consideration with respect to cable signals because of the greater power of such signals. Thus, according to one implementation, the impedance matching scheme for weak input signals (e.g. signals received by an antenna) may use active elements (e.g. transistors), which generate less noise than non-active elements (e.g. resistors). The impedance matching scheme when the input signals are sufficiently large (e.g. signals received by a cable or strong signals received by an antenna) may use non-active elements, which are compatible for use in a variable gain amplifier. Active element-based matching is difficult to implement in a broadband television signal amplifier having a variable gain. 
   In the exemplary implementation of the LNA  25  shown in  FIG. 6 , LNA  25  comprises first and second amplifiers  101  and  103  coupled to a switch  105 . The switch  105  functionally represents a mechanism that may be controlled to output a signal from one of the amplifiers  101  and  103 . According to one embodiment of the invention, the first amplifier  101  is adapted for the receipt of weak terrestrial signals and the second amplifier  103  is adapted for the receipt of cable and strong terrestrial signals. In particular, the first amplifier  101  may have a fixed gain, and may use active elements to perform impedance matching. The second amplifier  103  may have a variable gain, and may be configured such that impedance matching is implemented without using active elements. For example, the second amplifier  103  may implement impedance matching using a resistive ladder. 
   Switch  105  comprises first and second inputs  107  and  109  and an output  111 . The switch  105  further comprises a third input  113  for a switch control signal  181  to control the operation of the switch  105 . The control signal  181 , which may be generated by the AGC  31 , controls the selection of the first and second inputs  107 ,  109  to be connected to the output  111 . According to one embodiment of the invention, the first input  107  is coupled to the output  111  if the television signal received by the tuner has a power level below a given threshold, and the second input  109  is coupled to the output  111  if the television signal received by the tuner is has a power level above a given threshold. According to one exemplary implementation, the first input  107  is coupled to the output  111  if the power of the television signal received by the tuner is between approximately −85 dBm and −55 dBm, and the second input  109  is coupled to the output  111  if the power of the television signal received by the tuner is between approximately −55 dBm and +9 dBm. However, it should be appreciated that the invention is not limited in this respect. For example, control signal  181  may be generated in response to a manual switch that may be activated on the tuner, or in response to another stimulus. Terrestrial and cable television signals may be distinguished by power. Therefore, according to another exemplary implementation, the first input  107  is coupled to the output  111  if the television signal received by the tuner is a weak terrestrial television signal, and the second input  109  is coupled to the output  111  if the television signal received by the tuner is a cable or strong terrestrial television signal. 
   Preferably, the switch  105  is selected to allow for smooth switching between the first and second inputs  107 ,  109  so that discontinuities at the output  111  are lessened. According to one implementation, the switch  105  is a multiplexer. Discontinuities at output  111  may also be lessened by using currents at inputs  107  and  109  rather than voltages. To convert the voltage outputs of the amplifiers  101  and  103  to currents, transconductance amplifiers  115  and  117  may be coupled between the first and second amplifiers  101  and  103  and the first and second inputs  107  and  109 , respectively. 
   It should be appreciated that although the switch  105  is described above as being a single switch distinct from each of the first and second amplifiers  101  and  103 , the invention is not limited in this respect. For example, switch  105  may be implemented using a plurality of switches or other circuitry that enables multiplexing of signals. In addition, the one or more switches or circuitry that enables multiplexing may be integrated within the first and/or second amplifiers  101  and  103  themselves. Thus, it should be appreciated that switch  105  functionally represents a mechanism that enables switching between the first and second amplifiers  101  and  103 , but that the actual implementation of the switch need not be as illustrated in  FIG. 6 . 
   An exemplary implementation of a transconductance amplifier  137  such as the transconductance amplifiers  115 ,  117  of  FIG. 6  is shown in  FIG. 7 . Transconductance amplifier  137  comprises an input nodes  123   a–b  and output nodes  125   a–b . First and second transistors  127  and  129 , which may be bipolar junction transistors, are respectively coupled at their bases to the input nodes  123   a  and  123   b . The collectors of the first and second transistors  127  and  129  are respectively coupled to output nodes  125   a  and  125   b , into which flow currents, I 1  and I 2 . The emitters of the transistors  127 ,  129  are coupled by a resistor  131  having a resistance R. In addition, first and second current sources  133  and  135  are coupled between the emitters of the first and second transistors  127  and  129 , respectively, and ground. It should be appreciated that while the transconductance amplifier  137  illustrated in  FIG. 7  is shown as being implemented using bipolar devices, it is also possible to implement the transconductance amplifier using MOS devices. 
   The transconductance of the transconductance amplifier  137  may be expressed as follows, where V 1  is the voltage at the input node  123   a  and V 2  is the voltage at the input node  123   b : 
   
     
       
         
           Gm 
           = 
           
             
               
                 
                   I 
                   1 
                 
                 - 
                 
                   I 
                   2 
                 
               
               
                 
                   V 
                   1 
                 
                 - 
                 
                   V 
                   2 
                 
               
             
             = 
             
               1 
               R 
             
           
         
       
     
   
   Turning again to  FIG. 6 , a switch  119  may be used to prevent the first amplifier from consuming power when switch  105  is not selected to output the signal from the first amplifier  101 . The switch  119  may be located at the input of the first amplifier  101  and may be switchable to ground. The switch  119  may be controlled by the switch control signal  181  from AGC  31 , which also controls the state of switch  105 . It should be appreciated that although the switch  119  is illustrated as being a single switch distinct from each of the first and second amplifiers  101  and  103 , the invention is not limited in this respect. For example, switch  119  may be implemented using a plurality of switches or other circuitry, and may be integrated within the first and/or second amplifiers  101  and  103  themselves. Capacitor  121  may be located at the input of the first amplifier  101 , for example, for DC isolation of the first and second amplifiers  101 ,  103 . 
   As discussed above, the first amplifier  101  may be adapted for the reception of low power signals, which may be terrestrial signals or even weak cable signals, for example. In particular, the first amplifier  101  may have a fixed gain, and may use active elements to perform impedance matching. In one example, the input impedance of the first amplifier  101  is approximately 75 Ω. The use of active elements (e.g., transistors) to match the impedance of the terrestrial receiver may beneficially reduce noise. According to one example, the first amplifier may have a noise figure of approximately 5 dB or less. The gain of the first amplifier  101  may be selected based on the power of the terrestrial signals. In one example, the gain of the first amplifier is approximately 18 dB. 
   One exemplary implementation of the first amplifier  101  having a fixed gain is shown in  FIG. 8 . The broadband television signal is input to the amplifier  101  at input node  139  and the amplified signal is output at output node  141 . A transistor  143 , which may be a bipolar junction transistor, is coupled between the input node  139  and the output node  141 . Specifically, the base of transistor  143  is coupled to the input node  139 , and the collector of transistor  143  is coupled to the output node  141 . The first resistor  145  is coupled between the base and collector of transistor  143 . A second resistor  147  is coupled between a reference voltage V ref1  and the output node  141 . A third resistor  149  is coupled between the emitter of transistor  143  and ground. It should be appreciated that although the first amplifier  101  of  FIG. 8  is shown as having a single-ended configuration, the first amplifier  101  could alternatively be implemented differentially. 
   Where the transconductance (gm) of the transistor  143  is sufficiently high, the input impedance (Rin) of the first amplifier  101  may be described as follows, where the resistances of the first, second, and third resistors are R 1 , R 2 , and R 3 , respectively: 
             R   in     =         R   1     +     R   2         1   +       R   2       R   3                 
The gain of the first amplifier  101  may be described as follows:
 
           Gain   =       -     [         R   1     -     R   3         1   +       R   3       R   2           ]       *     [     1     R   in       ]             
In one exemplary implementation, R 1 , R 2 , and R 3  are selected such that the first amplifier  101  has an input impedance of 34.5 Ω and a gain of 18 dB. As discussed above, the second amplifier  103  may be adapted for the reception of higher power signals, which may be cable signals, for example. In particular, the second amplifier  103  may have a variable gain, and may use resistive elements to perform impedance matching. Resistive-based matching results in a higher noise figure than active element-based matching, but makes the variable gain implementation easier. The input impedance of the second amplifier  103  may be selected to match the impedance of a cable. The gain range of the second amplifier  103  may be selected based on the power of the cable signals. The gain range of the second amplifier  103  may have an upper value that is approximately the same as the gain of the first amplifier  101  to reduce the glitch when switching between the amplifiers. In one example, the gain of the second amplifier ranges from approximately −15 dB to approximately 18 dB. In another example, the gain range of the second amplifier is at least 18 dB. According to a further aspect of the invention, the gain of the second amplifier  103  may vary in response a gain control signal  179  generated by the AGC  31 . The gain control signal  179  may be generated in response to an indication of the power (e.g., the RMS or peak power) of the television signals received or processed. For example, a television signal  171  from the output of the IMF  29  may be processed to determine the power of the signal. As will be discussed in connection with  FIG. 9 , the second amplifier  103  may also receive a switch control signal  181 .
 
   One example of an amplifier that may be suitable for use as second amplifier  103  is the variable gain amplifier described in commonly assigned U.S. Pat. No. 5,077,541 to Gilbert, which is incorporated herein by reference. Another exemplary implementation of the second amplifier  103 , wherein the gain of the amplifier is variable, is shown in  FIG. 9 . A broadband television signal is input to the amplifier  103  at input node  151  and an amplified signal is output at output node  153 . The second amplifier  103  includes first, second, third, and fourth transconductance amplifiers  155   a–d , which may have transconductances that are independently variable. The output of each of the transconductance amplifiers  155   a–d  is connected to the output node  153  of the second amplifier  103 . First, second, and third resistors  157   a–c  are coupled between the inputs of the first, second, and third transconductance amplifiers  155   a–c , respectively, and a switch  159 . The switch is selectable between a bias voltage Vbias and ground. The switch  159  may be controlled by a control signal from AGC  31  ( FIG. 6 ) that controls the state of the switch  105  ( FIG. 6 ), such that the switch  159  is connected to bias voltage Vbias when the switch  105  ( FIG. 6 ) is selected to output the signal from the second amplifier  103  and connected to ground when it is not. Fourth, fifth, and sixth resistors  161   a–c  are respectively coupled between the inputs of the first and second transconductance amplifiers  155   a–b , the second and third transconductance amplifiers  155   b–c , and the third and fourth transconductance amplifiers  155   c–d . A seventh resistor  163  is coupled between the output node  153  of the second amplifier  103  and a reference voltage V ref2 . 
   The voltage Vo at the output node of the second amplifier  103  may be expressed by the following equation, where Gm 1 –Gm 4  are the transconductance values for the first through fourth transconductance amplifiers  155   a–d , Vin is the voltage at input node  151 , and R 163  is the resistance of resistor  163 : 
             V   o     =       [       [       V   in     *     Gm   4       ]     +     [         V   in     2     *     Gm   3       ]     +     [         V   in     4     *     Gm   2       ]     +     [         V   in     8     *     Gm   1       ]       ]     ⁢     R   163             
Input  167  may be used to control the transconductance values Gm 1 –Gm 4 . According to one example shown in  FIG. 6 , AGC  31  may supply a gain control signal  179  to the second amplifier  103  based on a desired gain of the second amplifier  103 . The AGC  31  may select the desired gain based on a detected power of a television signal  171  at the output of the IMF  29 . Gain control signal  179  may be provided to the transconductance amplifiers  155   a–d  via the input  167  of the second amplifier  103 .
 
   According to one exemplary implementation, each of transconductance amplifiers  155   a–d  may be implemented as shown in  FIG. 7 , but where resistor  131  is provided with a variable resistance. Varying the resistance of the resistor  131  changes the transconductance of the amplifier. 
   An exemplary implementation of the AGC  31  of  FIG. 6  is illustrated in  FIG. 10 . The AGC  31  receives a television signal  171 , which may be received from the output of the IMF  29  as shown in  FIG. 6 . The television signal may alternatively be received from another source, such as the input of the tuner. A power detector  169  measures the power of television signal  171 . In one implementation, the power detector includes an envelope detector to generate the average envelope of the broadband signal from which an average power of the signal can be determined. An analog-to-digital converter (ADC)  173  then digitizes the output of the power detector  169  and outputs signal  174 . The output of the ADC  173  and a reference power value  175  are input to a controller  177 . The reference power value  175  may represent the desired peak or RMS power of the television signal at the output of the IMF  29  ( FIG. 6 ). According to one example, the controller  177  may be a proportional derivative (PD) controller. Controller  177  generates a gain control signal  179  based on the output of the ADC  173 , signal  174 , and the reference power value  175 . The gain control signal  179  is supplied to the second amplifier  103  via input  167  and used to program the transconductance of each of the transconductance amplifiers  155   a–d . Each of the transconductance amplifiers  155   a–d  may separately receive a different input (i.e., some number of bits) derived from the output of the controller  177 . 
   Another controller, controller  180 , may generate a switch control signal  181  to control activation of switch  105 , as shown in  FIG. 6 . In addition, the switch control signal  181  may be sent to each of first and second amplifiers  101  and  103 , shown in  FIG. 6 , and may be used to disable the amplifier that is not connected to output  111  by switch  105 . As discussed previously, switch control signal  181  may be based on the detected power of a broadband television signal (e.g., television signal  171 ), in connection with an embodiment of the invention. According to one implementation illustrated in  FIG. 10 , controller  180  may be used to generate switch control signal  181  in response to a detected power of television signal  171 . 
   Controller  180  receives signal  174 , which is a digital signal representing the power of television signal  171 , and outputs switch control signal  181 . The switch control signal  181  may be generated by the controller  180  based on hysteresis curve  235 , such as that shown in  FIG. 11 . Hysteresis curve  235  illustrates the switch control signal  181  as a function of signal  174 . The hysteresis curve  235  comprises two thresholds, one at −43 dBm and one at −40 dBm. When signal  174  is increasing, switch control signal  181  increases from a first level  237  to a second level  239  when signal  174  increases past −40 dBm. When signal  174  is decreasing, switch control signal  181  decreases from the second level  239  to the first level  237  when signal  174  decreases past −43 dBm. When switch control signal  181  is at the first level  237 , amplifier  101  is activated and amplifier  103  is unactivated. When switch control signal  181  is at the second level  239 , amplifier  103  is activated and amplifier  101  is unactivated. The use of different thresholds for switching amplifiers depending on whether power is increasing or decreasing may reduce oscillation. It should be appreciated that the hysteresis curve  235  shown in  FIG. 11  is merely exemplary, and that other thresholds or functions for controlling switch control signal  181  may be used. 
   An exemplary implementation of the image filter (IMF)  29  of  FIG. 6  is shown in  FIG. 12 . The IMF  29  includes input nodes  183   a–b  and output nodes  185   a–b . The output nodes  185   a  and  185   b  are respectively coupled between the emitters of first and second transistors  187  and  189  and one end of the first and second current sources  191  and  193 . The other end of each of the first and second current sources  191  and  193  is coupled to ground. The bases of transistors  187  and  189  are respectively coupled to one end of capacitors  195  and  197  and inductors  199  and  201 . The other end of each of capacitors  195  and  197  is coupled to a reference voltage V ref . The other ends of inductors  199  and  201  are respectively coupled to resistors  203  and  205  at input nodes  183   a  and  183   b . An end of each resistors  203  and  205  is also coupled to the reference voltage V ref , as is the collector of each of transistors  187  and  189 . It should be appreciated that while the IMF  29  illustrated in  FIG. 12  is shown as being implemented using bipolar devices, it is also possible to implement the IMF using MOS devices. 
   The quality factor (Q) of the IMF  29  may be expressed by the following equation: 
           Q   =       1   R     ⁢       L   C               
where R represents the resistance of resistors  203  and  205 , L represents the inductance of the inductors  199  and  201 , and C represents the capacitance of capacitors  195  and  197 . Similarly, the natural frequency (Ω 0 ) of the IMF  29  may be expressed by the following equation:
 
             Ω   o     =     1     LC             
where L represents the inductance of the inductors  199  and  201  and C represents the capacitance of capacitors  195  and  197 .
 
   Having described several illustrative embodiments of the invention, various alterations, modifications and improvements will readily occur to those skilled in the art. Such alterations, modifications and improvements are intended to be in the spirit and scope of the invention. Accordingly, the foregoing description is by way of example only and is not intended as limiting. The invention is limited only as defined in the following claims and the equivalents thereto.