Abstract:
A bipolar/CMOS mixed type switching circuit comprising two npn-type bipolar transistors Q 1 , Q 2  that are connected in the form of a totem pole in the output stage, a CMOS inverter and an NMOSFET M 3  for driving these transistors in a complementary manner, and resistance means R for discharging the electric charge stored in the base of the transistor Q 2 . The threshold voltage of an NMOSFET M 2  constituting the CMOS inverter in the absence of substrate effect is set to be substantially equal to the threshold voltage of the NMOSFET M 3  in the absence of the substrate effect, and the channel conductance W N  /L N  of the NMOSFET M 3  is so set that the threshold voltage V LT1  of the CMOS inverter and the practical threshold voltage V LT2  of the NMOSFET M 3  will be nearly the same. Owing to the above structure, there is obtained a switching circuit which permits little through current to flow and which operates at high speeds.

Description:
BACKGROUND OF THE INVENTION 
     The present invention relates to a switching circuit. 
     A composite technique has been developed to form different kinds of elements in a single semiconductor substrate, to obtain a semiconductor integrated circuit having diverse functions and a high degree of integration. 
     For instance, a circuit technique for combining bipolar transistors with insulated gate-type fieldeffect transistors (hereinafter referred to as MOSFET&#39;s) has been disclosed in Japanese Patent Publication No. 43997/1972 and in Japanese Patent Laid-Open No. 26181/1977. 
     FIG. 1 shows a switching circuit which was contrived by the inventors of the present invention and in which a bipolar transistor and an insulated gate-type field effect transistor are combined. (See U.S. Pat. Ser. No. 513,056 which is hereby incorporated by reference.) The circuit shown in FIG. 1 is an input buffer circuit (switching circuit) used for, for example, in a Bi-CMOS (bipolar/CMOS mixed type) gate array. The circuit consists of two bipolar transistors Q1, Q2 that constitute an output stage, a CMOS inverter 12 which drives the bipolar transistor Q1 in an inverted manner, and a buffer amplifier (voltage follower) 14 which drives the other bipolar transistor Q2 in a non-inverted manner. 
     This circuit operates as described below. A logic signal applied to an input terminal IN is divided into two branches. One part of the input is phase inverted by the CMOS inverter 12 and is input to the base of the transistor Q1 of the output stage. The other part of the input is converted into a low impedance by the buffer amplifier 14 and is input in phase to the base of the other bipolar transistor Q2 of the output stage. Therefore, the two bipolar transistors Q1, Q2 in the output stage are rendered conductive and are driven in a complementary manner. When one transistor Q1 is ON (conductive) and the other transistor Q2 is OFF (nonconductive), a changing current is supplied to the load Co through the transistor Q1. When one transistor Q1 is OFF and the other transistor Q2 is ON, the electric charge stored in the load Co is discharged through the other transistor Q2. Accordingly, the capacitive load Co is driven in this fashion. 
     The switching circuit has the features (effects) described below. 
     (1) The CMOS inverter 12 and the buffer amplifier circuit 14 have nearly the same signal transmission speed; hence, the bases of the two transistors Q1, Q2 are driven nearly at the same timing in an opposite phase relation. Therefore, the two transistors Q1, Q2 are turned on simultaneously for only a short time, making it possible to decrease the through current. 
     (2) The two transistors Q1, Q2, which are of the npn-type, can be used to constitute the output stage. When a semiconductor integrated circuit is constructed, therefore, a high cut-off frequency f T  can be easily obtained to realize a high operation speed. 
     (3) When the bipolar transistor Q1 in the output stage is turned off, the electric charge accumulated in the base thereof can be quickly extracted through a MOSFET M2 of the CMOS inverter 12. When the other bipolar transistor Q2 in the output stage is turned off, the electric charge accumulated in the base thereof can be quickly extracted by a low-impedance output of the voltage follower 14. That is, the two bipolar transistors Q1, Q2 in the output stage, respectively, have paths for effectively extracting the electric charge accumulated in the bases. Therefore, the switching time from ON to OFF is conspicously shortened. 
     (4) Since a power source terminal p1 of the voltage follower 14 is connected to the output terminal OUT, the discharging current of the capacitive load Co connected to the output terminal OUT flows not only to the other transistor Q2 in the output stage but also to the voltage follower 14 as an operation current from the first power source terminal p1. At the moment when the logic state of the buffer output OUT changes from &#34;H&#34; (high logic level) to &#34;L&#34; (low logic level), the electric charge stored in the load Co is allowed to discharge through the transistor Q2 and the voltage follower 14. Therefore, the driving power is greatly reinforced for the capacitive load Co, especially at the moment of breaking. 
     (5) Further, since the CMOS inverter 12 and the voltage follower 14 have high input impedances, there is obtained a high input impedance as viewed from the input side. 
     (6) The first power source terminal p1 of the voltage follower 14 is connected not to the power source V DD  but to the collector (output terminal OUT) of the transistor Q2 of the output stage, and the base potential of the transistor Q2 is not higher than the collector potential thereof. Therefore, the transistor Q2 is not saturated. 
     The switching circuit exhibits excellent features as described above. Further study of the problem enabled the inventors to realize the switching circuit in the form of an integrated circuit. They have found that in designing constants for the circuit, many contrivances are necessary to satisfy the high-speed characteristics and the low power consumption that are strictly necessary for the switching circuit. The present invention was achieved through the above study. 
     SUMMARY OF THE INVENTION 
     The object of the present invention is to provide a novel switching circuit having high performance, that can be suitably realized in the form of a semi-conductor integrated circuit. 
     The switching circuit is constructed as described below. 
     (1) The circuit comprises two npn-type bipolar transistors Q 1 , Q 2  in the output stage connected in the form of a totem pole, CMOS inverters M 1 , M 2  for driving the base of the transistor Q 1 , a source follower M 3  for driving the base of the transistor Q 2 , a resistor R of which the one end is commonly connected to the base of the transistor Q 2  and to the source of the source follower M 3 , and an input signal terminal commonly connected to the gate of the source follower M 3  and to the gate of the CMOS inverters M 1 , M 2 . 
     (2) A threshold voltage V thNO  of n-channel MOSFET&#39;s constituting the CMOS inverters is selected to be substantially equal to a threshold voltage V thNO  of the MOSFET (source follower) M 3 . Here, the threshold voltage V thNO  stands for that of the n-channel MOSFET when there is no substrate effect. 
     (3) Resistance of the resistor R is selected to lie over a predetermined range in order to set the turn-on time and the turn-off time of the NPN bipolar transistor Q 2  to be shorter than a predetermined value. 
     (4) The channel conductance Wn/Ln of the source follower M 3  has been so set that the threshold voltage V LT2  of the source follower M 3  will be close to the threshold voltage V LT1  of the CMOS inverters. Here, Ln denotes gate length, and Wn denotes gate width. 
     Owing to the above-mentioned structure, a high-speed switching circuit which permits little through current to flow can be obtained without increasing the complexity of IC manufacturing. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a circuit diagram showing fundamental circuit structure of the switching circuit that serves as a prerequisite of the present invention; 
     FIG. 2 shows structure of the switching circuit according to the first embodiment of the present invention; 
     FIG. 3 is a diagram showing the relation between the resistance of the resistor R and the threshold voltage V LT2  of the source follower circuit; 
     FIG. 4 is a diagram showing the relation between the resistance of the resistor R and the turn-on and turn-off times of the bipolar transistor Q2; 
     FIG. 5 is a circuit diagram illustrating a second embodiment of the present invention; 
     FIG. 6 is a circuit diagram illustrating a third embodiment of the present invention; and 
     FIG. 7 is a circuit diagram illustrating a fourth embodiment of the present invention. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Embodiment 1 
     FIG. 2 is a circuit diagram which concretely illustrates a first embodiment of the present invention. 
     The present invention was achieved through the study of how to reduce the through current of the switching circuit, how to increase the signal transmission speed, and how to realize the switching circuit in the form of a semiconductor integrated circuit. 
     Therefore, the progress of study by the inventors will first be described below, and the features of the present invention will then be described. 
     The circuit shown in FIG. 2 was studied by the inventors through the process as mentioned below. 
     I. Consideration of logic threshold voltage V LT1  of the CMOS inverter 12: 
     The substrate effect takes place if the source S of NMOS-FET M 2  is connected to the base of the transistor Q 2  as indicated by a solid line. Here, the term &#34;substrate effect&#34; means that if the source potential so changes as to become higher than or lower than the ground potential while the potential of the silicon substrate has been fixed to ground potential, the practical threshold voltage of the MOSFET changes depending upon the change in the source potential. If the substrate effect is taken into consideration, therefore, the threshold voltage V thN  of NMOSFET M 2  in practice is given by the following well-known equation: ##EQU1## where V thNO  denotes threshold voltage when there is is no substrate effect, 
     ΔV th  denotes shifted amount caused by the substrate effect, 
     K denotes a constant, 
     2φ F  denotes voltage that is twice as great as the Fermi potential φ F , 
     V BS  denotes voltage across the substrate and the source of the NMOSFET M 2 , and 
     V BE  denotes voltage across the base and the emitter of the transistor Q 2 . 
     Next, ##EQU2## is defined below. 
     In the above equation, β P  and β N  denote conductance (constant) of the p-channel MOSFET and conductance of the n-channel MOSFET, respectively, and β PO  and βNO denote values of β P  and β N , respectively, when W/L=1 (where W is gate width and L is gate length). 
     Then, a logic threshold voltage V LT1  of the CMOS inverter 12 is given by, ##EQU3## where V DD  is power source voltage, and 
     V thPO  denotes the threshold voltage of PMOSFET M 1  when there is no substrate effect. 
     Generally, α is set to a suitable value to so design the circuit that V LT1  ≈1/2V DD  (note here that the symbol ≈ is used to indicate &#34;substantially equal to&#34;). Here, the threshold voltage V LTl  of the CMOS inverter stands for a gate voltage applied to the CMOS inverter when an electric current flows substantially equally into the PMOSFET M 1  and NMOSFET M 2  that constitute the CMOS inverter. 
     II. Consideration of the logic threshold voltage V LT2  of the NMOS source follower circuit: 
     It is assumed for purposes of beginning the analysis that the logic threshold voltage V LT2  for turning the NMOSFET M 3  and the transistor Q 2  from off to on, is ##EQU4## 
     III. Consideration of through currents of the transistors Q 1 , Q 2  : 
     To prevent the flow of through currents of the transistors Q 1 , Q 2 , the relation V LT1  ≈V LT2  must hold true. If V DD  =5 volts, V LT1  ≈2.5 volts. In order for the logic threshold voltage V LT2  to be 2.5 volts when V BE  =0.6 volt, it is necessary to implant impurity ions into the channel region of the NMOSFET M 3 , so that V thNO  will become 1.9 volts. Here, V thPO  of M 1  is -0.6 volt and V thNO  of M 2  is +0.6 volt. To set the threshold voltage V thNO  of NMOSFET M 3  to 1.9 volts, therefore, the individual MOSFET&#39;s, M 2  and M 3  must be formed through individual processes, or must be formed through processes that are partly common to each other; then, the desired threshold voltages of each of the MOSFET&#39;s must be obtained through additional processes. 
     IV. Consideration of the NMOS source follower M 3  and the logic threshold voltage V LT2  of the transistor Q 2  : 
     The inventors have furthered the study concerning the logic threshold voltage V LT2 , and have found the fact that the logic threshold voltage V LT2  is not simply found from the equation (3) above but varies with the resistance of the resistor R as well as β NO  of NMOSFET M 3  and W N  /L N  of NMOSFET M 3 , as shown in FIG. 3. 
     The reason why the logic threshold voltage V LT2  varies according to the relation shown in FIG. 3 will be analyzed below. 
     Here, if ##EQU5## the current that flows through the drain-source path of the NMOSFET M 3  is: ##EQU6## where V GS  denotes the voltage across the gate and the source of NMOSFET M 3 . 
     Input voltage V IN  at the input terminal IN is: 
     
         V.sub.IN =V.sub.GS +R·I.sub.DS                    (5) 
    
     The transistor Q 2  is rendered conductive when the voltage drop R·I DS  of the resistor R satisfies the following equation: 
     
         V.sub.BEQ.sbsb.2 =R·I.sub.DS                      (6) 
    
     From equations (5), (6), there is obtained, 
     
         V.sub.GS =V.sub.IN -V.sub.BE                               (7) 
    
     From equations (4), (7), there is further obtained, ##EQU7## 
     If both sides of the equation (8) are multiplied by R, and if V BE  =R·I DS  is taken into consideration, there is obtained the following equation: ##EQU8## 
     By modifying equation (9), there is obtained, ##EQU9## 
     By modifying equation (10), there is further obtained, ##EQU10## 
     From equation (12), it will be understood that the logic threshold voltage V LT2  also varies depending upon β NO , W N  /L N  and R. FIG. 3 shows the relation between the resistance of the resistor R and the threshold voltage V LT2  of the NMOSFET M 3  relying upon the results measured according to the present invention, wherein a solid line represents the relation when W N  /L N  =10μm/2μm, a dot-dash chain line represents the relation when W N  /L N  =20/2, and a two-dot chain line represents the relation when W N  /L N  =40/2. 
     V. Consideration of the relation between the resistance of the resistor R and turn-on time t ON  and turn-off time t OFF  of the transistor Q 2  : 
     FIG. 4 shows the relation between the resistance of the resistor R and t ON , t OFF , that is practically examined. 
     In FIG. 4, the solid line represents the turn-on time t ON , and a dotted line represents the turn-off time t OFF . 
     The following facts will be understood from FIG. 4. 
     (1) To set t ON  and t OFF  to be shorter than 2 nsec., the resistance must be selected to be 1KΩ&lt;R&lt;30KΩ (range A). 
     (2) To set t ON  and t OFF  to be shorter than 1.5 nsec., the resistance must be selected to be 3KΩ&lt;R&lt;20KΩ (range B). 
     (3) To set t ON  and t OFF  to be shorter than 1.25 nsec., the resistance must be selected to be 4KΩ&lt;R&lt;16KΩ (range C). 
     In the foregoing were mentioned the results studied by the inventors. The specific construction of the circuit shown in FIG. 2 will further be described below. 
     In FIG. 2, the source S of the n-channel MOSFET M 2  may be either grounded as indicated by a dotted line or be connected to the base of the transistor Q 2  as indicated by a solid line. When the source of the NMOSFET M 2  is connected to the base of the transistor Q 2 , however, it becomes difficult to design the threshold voltage V LT1 , of the CMOS inverter 12. When the source of the NMOSFET M 2  is grounded, it is easy to design the threshold voltage V LT1 . 
     The circuit of FIG. 2 is designed in four steps: 
     Step 1: The threshold voltage V thNO  of NMOSFET M 2  is set to be substantially equal to the threshold voltage V thNO  of NMOSFET M 3 . For instance, NMOSFET M 2 , M 3  are formed on the same chip by the same manufacturing process. 
     Step 2: The resistance R is set to lie within a predetermined range to set t ON  and t OFF  of the transistor Q 2  to be shorter than a predetermined value. For instance, to set t ON , t OFF  to be shorter than 2 nsec., the resistance R is selected to lie within the aforementioend range A. 
     Step 3: Design the threshold voltage V LT1  of the CMOS inverter which consists of PMOSFET M 1  and NMOSFET M 2 . That is, when the source of the NMOSFET M 2  has been grounded, the constant of the parameter is determined as: ##EQU11## when the source of the NMOSFET M 2  has been connected to the base of the transistor Q 2 , the threshold voltage should be designed in accordance with the equation (2) mentioned earlier. 
     Step 4: Resistance R over the range mentioned in Step 2 is used so that the threshold voltage V LT2  of the NMOSFET M 3  will approach the threshold voltage V LT1  that is set in Step 3, and values β NO , W N  /L N  are set in accordance with the equation (12), such that V LT1  ≈V LT2 . Here, however, β NO  serves as a constant that does not change once the manufacturing process is determined. In practice, therefore, V LT1  ≈V LT2  is accomplished by changing W N  /L N  of NMOSFET M 3 . 
     Described below is a concrete example when the above-mentioned design procedure is followed (where the source of NMOSFET M 2  is grounded in FIG. 2). 
     Step 1: Set V thNO  of NMOSFET&#39;s M 2  and M 3  to be equal to each other. 
     Step 2: Set the resistance R to be 8KΩ so that the turn-on and turn-off times of the transistor Q 2  will be shorter than 1.25 nsec. 
     Step 3: Set the value W P  /L P  of PMOSFET M 1  constituting the CMOS inverter to be 30/2, and set the value W N  /L N  of NMOSFET M 2  to be 10/2. 
     Here, ##EQU12## 
     Therefore, ##EQU13## 
     Step 4: From the equation (12), there is obtained, ##EQU14## 
     Since V LTl  =2.5 V≈V LT2 , if 
     V LT2  =2.5 V 
     V BE  =0.6 V 
     V thN  =0.6 V 
     are inserted into the equation (13), there is obtained, ##EQU15## If W N  /L N  is found from the equation (14), we obtain W N  /L N  ≈5/2. That is, the value W N  /L N  of NMOSFET M 3  should be set to 5/2. 
     The above-mentioned structure makes it possible to obtain the following effects in addition to the effects mentioned in the paragraph of Background of the Invention. 
     (1) The threshold voltage V thNO  (when there is no substrate effect) of NMOSFET M 2  constituting the CMOS inverter 12 is set to be substantially equal to the threshold voltage V thNO  (when there is no substrate effect) of NMOSFET M 3 . This means that the NMOSFET&#39;s M 2  and M 3  can be formed simultaneously in the semiconductor substrate through the same manufacturing process, thereby simplifying the manufacture of integrated circuits. 
     (2) Resistance of the resistor R is so determined that the bipolar transistor Q 2  in the output stage exhibits a turn-on time and a turn-off time of desired values (high speeds). Therefore, the high switching speed of the transistor Q 2  is correctly determined. 
     (3) Value W N  /L N  of the NMOSFET M 3  is so adjusted that the logic threshold voltage V LT2  of NMOSFET M 3  approaches the logic threshold voltage V LT1  of the CMOS inverter. Therefore, the two transistors Q 1   and Q 2  in the output stage are driven at nearly the same timing in a complementary manner, making it possible to minimize the through current that flows instantaneously through the transistors Q 1 , Q 2 . 
     Embodiment 2 
     FIG. 5 shows the structure of a switching circuit according to a second embodiment of the present invention. What makes this embodiment different from the circuit structure of embodiment 1 is that the resistor R in this embodiment is formed by utilizing the resistance of a MOSFET M 4  while it is conductive. 
     Similar to embodiment 1, the circuit in this embodiment is designed in five steps. 
     Step 1: The threshold voltage V thNO  of NMOSFET M 2  is set to be substantially equal to the threshold voltage V thNO  of NMOSFET M 3   
     Step 2: To set t ON  of the transistor Q 2  (time required for turning Q 2  from off to on) to be shorter than a predetermined value, the resistance R of NMOSFET M 4  while it is conductive is set to lie within a predetermined range. 
     To set t ON  to be shorter than 2 nsec., R must be greater than 1KΩ. 
     To set t ON  to be shorter than 1.5 nsec., R must be greater than 3 KΩ. 
     To set t ON  to be shorter than 1.25 nsec., R must be greater than 4KΩ. 
     Step 3: In order to set t OFF  required for turning the transistor Q 2  from on to off to be shorter than a predetermined value, the resistance R of NMOSFET M 4  when it is turned from off to on is set to lie within a predetermined range, the NMOSFET M 4  being driven by a current that flows through a path consisting of input terminal IN, PMOSFET M 1 , and transistor Q 1 . 
     To set t OFF  to be shorter than 2 nsec., R≦30 KΩ. 
     To set t OFF  to be shorter than 1.5 nsec., R&lt;20KΩ. 
     To set t OFF  to be shorter than 1.25 nsec., R&lt;16KΩ. 
     Step 4: Design the threshold voltage V LT1  of the CMOS inverter consisting of M 1  and M 2 . When the source of NMOSFET M 2  is grounded, design the threshold voltage according to the following equation, ##EQU16## When the source of NMOSFET M 2  is connected to the base of the transistor Q 2 , design the threshold voltage according to the aforementioned equation (2). 
     Step 5: Use the resistance R of NMOSFET M 4  over the ranges of Steps 2 and 3, that V LT2  will approach V LT1  that has been set in Step 4, and set β NO  and W N  /L N  in accordance with the equation (12) to accomplish the relation V LT1  ≈V LT2 . 
     Embodiment 3 
     FIG. 6 is a diagram showing a switching circuit according to a third embodiment of the present invention. 
     The feature of this circuit resides in the provision of a collector-grounded pnp-type bipolar transistor Q 3  in the input portion. 
     If the circuit is based on the prerequisite that an input signal (high level V 1  H=2.0 V, threshold level V Ith  =1.3 V, low level V 1  L=0.8 V) of the TTL level is applied to the input terminal IN, the threshold voltage V Ith  of the transistor Q 3  must be set to 1.3 volts. In this case, the design should be carried out according to Steps 1 to 4, such that V LT1  =V LT2  =V Ith  +V BE  =1.3 V+0.6 V=1.9 volts. 
     Embodiment 4 
     According to this embodiment, shown in FIG. 4, the switching circuit is provided with a NOR logic function for a pair of input signals IN A  and IN B  relying upon a plurality of MOSFET&#39;s M1A, M1B, M2A, M2B, M3A and M3B. As can be seen in FIG. 7, the input IN A  is coupled to the gates of M1A, M2A and M3A while the input IN B  is coupled to the gates of M1B, M2B and M3B for similarly producing an output. By virtue of the series connection of M1A and M1B and the parallel connections of M2A and M2B as well as M3A and M3B, a NOR output at OUT will be produced for the inputs IN A  and IN B . The transistor M 4  serves as the resistor R in this circuit in a manner similar to FIG. 5. 
     The design procedure for setting values for the transistors and the resistance is the same as the one described above. 
     Although particular values for voltages, resistances and dimensions have been provided in the foregoing description, it is to be understood that these are for purposes of example in conjunction with the described embodiments, and the present invention is not necessarily limited to such values. On the contrary, the principles and steps provided in the foregoing description can be used to design switching circuits having reduced through current for a variety of different values. 
     Also, although the invention is described in terms of MOSFET&#39;s (technically meaning Metal-Oxide-Semiconductor FETs), it is to be understood that this is done in the more general meaning now attached to this term, which includes other IGFETs (Insulated-Gate FETs) which may have their gates formed of material other than metal (e.g. doped polycrystalline silicon) and their gate insulation formed of material other than oxide (e.g. Si 3  N 4 ). 
     Further, although the fourth embodiment has been provided to show connection of the elements of the present invention in a logic NOR configuration, it is to be understood that the circuit could be arranged to provide other logic functions, if desired, while still operating with the CMOS inverter arrangement and source follower arrangement discussed for this invention. 
     It is to be understood that the above-described arrangements are simply illustrative of the application of the principles of this invention. Numerous other arrangements may be readily devised by those skilled in the art which embody the principles of the invention and fall within its spirit and scope.