Abstract:
A method is disclosed for operating a radio frequency transmitter chain component, as is a radio frequency transmitter chain component that operates in accordance with the method. The method includes receiving an input signal to be mixed with a signal output from an oscillator, where the input signal is received through an operational amplifier. The method further includes applying an output of the operational amplifier to an input of a mixing circuit, rectifying the input signal to produce a rectified input signal and controlling a common-mode output voltage of the operational amplifier with the rectified input signal. This process varies the power consumption of the component in a manner that is proportional to a value of the input signal. A further step couples a mixer output signal to an input of a VGA. The component may include both the mixer and the VGA. In the preferred embodiment the input signal and the mixer output signal are differential signals.

Description:
TECHNICAL FIELD 
   These teachings relate generally to radio frequency (RF) circuitry and, more specifically, to transmitters of portable radiocommunication terminals, also referred to here in as mobile stations. 
   BACKGROUND 
   Referring to  FIG. 1 , the basic architecture of a conventional RF transmitter chain  1  includes a digital-to-analog converter (DAC)  2 , a filter  3 , a mixer (up-conversion to the transmitted frequency)  4 , a gain control circuit (Variable Gain Amplifier or VGA)  5  and a power amplifier  6  having an output coupled to a transmit antenna  7 . The VGA  5  is used to adjust the output power of the transmitter chain  1  to the desired level. 
   In the conventional transmitter chain  1  the current consumption is typically constant regardless of the value of the input signal, typically represented by In-phase (I) and Quadrature (Q) input signal channels. However, there should be no signal at the transmitter output if the IQ input signal is zero. This condition implies, ideally, that the transmitter current consumption could then also be zero. Prior to this invention, however, it was not possible to adjust the current consumption of the transmitter chain such that only a required amount of current is consumed with each IQ input value. 
   Said another way, a change in the transmitter bias signal, and hence a change in current consumption, was not possible with prior art transmitter topologies, since the change in the transmitter bias signal would result in an undesirable amplitude modulation of the transmitted carrier signal. More specifically, by changing the current consumption of an amplifier the gain of the amplifier is also changed. Therefore, if one attempted to change the bias to, and hence, the current consumption of the mixer  4 , the gain would also change. The change in the amplifier gain resulted in the introduction of the undesirable amplitude modulation of the transmitter output signal. 
   Reference can be had to the following commonly assigned U.S. Patents for descriptions of various prior art VGA circuits and techniques, used in RF transmitters as well as in RF receivers: U.S. Pat. No. 5,548,616, L. Mucke et al.; U.S. Pat. No. 5,752,172, J. Matero; U.S. Pat. No. 5,752,170, P. Clifford; U.S. Pat. No. 5,884,149, M. Jaakola; U.S. Pat. No. 6,009,129, T. Kenney et al; U.S. Pat. No. 6,060,950, J. Groe; U.S. Pat. No. 6,167,273, G. Mandyam; U.S. Pat. No. 6,084,471, R. Ruth, Jr. et al.; U.S. Pat. No. 6,178,313 B1, P. Mages et al.; U.S. Pat. No. 6,317,589 B1, A. Nash and U.S. Pat. No. 6,370,358 B2, J. Liimatainen. 
   SUMMARY OF THE INVENTION 
   The foregoing and other problems are overcome by methods and apparatus in accordance with embodiments of this invention. 
   In one aspect this invention provides a method for operating a radio frequency transmitter chain component, and a radio frequency transmitter chain component that operates in accordance with the method. The method includes receiving an input signal to be mixed with a signal output from an oscillator, where the input signal is received through an operational amplifier. The method further includes applying an output of the operational amplifier to an input of a mixing circuit, rectifying the input signal to produce a rectified input signal and controlling a common-mode output voltage of the operational amplifier with the rectified input signal in order to vary the power consumption of the component in a manner that is proportional to a value of the input signal. A further step couples a mixer output signal to an input of a VGA. In the preferred, but non-limiting embodiment, the component comprises both the mixer and the VGA. In the preferred embodiment the input signal and the mixer output signal are differential signals. 
   In a further aspect this invention provides a radio frequency transmitter chain component that includes a mixer for receiving a differential input signal to be mixed with a signal output from an oscillator, where the component includes an input differential operational amplifier for receiving the input signal and for outputting differential signals to a Gilbert cell mixing circuit. The operational amplifier operates with common-mode feedback set by a rectified version of the input signal. The component may further include a VGA having differential inputs coupled to outputs of the Gilbert cell mixing circuit. In the preferred embodiment the input signal comprises one of an Inphase (I) or a Quadrature (Q) analog signal. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The above set forth and other features of these teachings are made more apparent in the ensuing Detailed Description of the Preferred Embodiments when read in conjunction with the attached Drawings, wherein: 
       FIG. 1  is a simplified block diagram of a conventional RF transmitter chain; 
       FIG. 2  is a simplified block diagram of an RF transmitter chain having a combined MIXER/VGA circuit in accordance with this invention; 
       FIG. 3  is a schematic diagram, partly in block diagram form, of the MIXER/VGA circuit shown in  FIG. 2 ; and 
       FIG. 4  is a schematic diagram of the I-Mixer block shown in  FIG. 3 . 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   In the prior art transmitter  1  the mixer  4  and the VGA  5  were separate components, and by changing the current consumption of the amplifier the gain of amplifier was also changed. As a result, the gain of the mixer  4  and the VGA  5  would vary if their current consumption were altered. 
   Referring to  FIG. 2 , in the presently preferred embodiment of this invention the mixer  4  and the VGA  5  are combined into one component, shown as a MIXER/VGA  10 , and a change in the current consumption is achieved as a function of the input signal, i.e., the IQ input values to the DAC  2 . This results in an improved transmitter chain  1 A of a mobile station  100 , such as a cellular telephone or a personal communicator, as compared to the convention transmitter chain  1 . 
   Referring also to  FIGS. 3 and 4 , in the preferred embodiment the current consumption is varied as a result of feedback generated by the inclusion of an operational amplifier (opamp)  12  at the input of the mixer. The presence of the opamp  12  reduces the gain variation of the MIXER/VGA  10 , due to a change in bias, by the open-loop gain of the opamp  12 . The feedback signal is generated from the output of the opamp  12  where it drives a mixer circuit that, in the presently preferred embodiment, is constructed as a Gilbert cell (see  FIG. 4 ). 
   In  FIG. 2  the DAC  2  generates analog IQ-signals  2 A and  2 B. The analog IQ signals  2 A and  2 B are low pass filtered in the filter  3 , and fed as filtered analog IQ signals  3 A and  3 B to the combined MIXER/VGA  10  that both mixes the signal to the higher (carrier) frequency (e.g., about 2 Ghz) and adjusts the output swing, i.e., controls the gain of the modulated carrier signal. The gain controlled carrier signal is then fed to the power amplifier  6 , which feeds the signal to the antenna  7  with constant gain. As examples, in a wideband code division, multiple access (WCDMA) embodiment the bandwidth of the analog IQ signals  3 A and  3 B is approximately 2.5 MHz, while in a GSM embodiment the bandwidth is approximately 150 kHz. 
     FIG. 3  shows the MIXER/VGA  10  to be constructed of an I-Mixer  14  and a Q-Mixer  16  having outputs that feed a transistor quad  18  (transistors T 1 , T 2 , T 3  and T 4 ). A quad control voltage controls the gain of the quad  18 , and thus the gain of the VGA portion  10 A of the MIXER/VGA  10 . The quad  18  is composed of bipolar junction transistor (bjt) differential pairs and forms the gain control element of the VGA  10 A. The quad control voltage (gain control) is typically applied from a control device, such as a DSP, through a temperature compensation block (not shown), and causes the gain of the VGA  10 A to change. Current is steered between the bipolar junction transistors (bjts) of the quad  5 A according to the quad (gain) control voltage. 
     FIG. 4  shows a schematic diagram of the I-Mixer  14  of  FIG. 3 , it being realized that the Q-Mixer  16  is identically constructed, although the signal designations are changed (e.g., IP=QP, IM=QM, LOP_I=LOP_Q, etc.). In  FIG. 4  the signals LOP_I and LOM_I are input signals from a local oscillator (LO), typically a voltage controlled oscillator (VCO, not shown), and represent the frequency signals to be mixed with IP and IM. 
   The gain of the PA  6  is typically about 20 dB to 40 dB, and the required output power is typically in excess of 20 dBm. As a result, the required output power from the MIXER/VGA  10  is about 0 dB. 
   In accordance with this invention the varying I and Q voltages  3 A and  3 B are rectified to DC, preferably by full-wave rectifiers (FWRs)  22 A and  22 B shown in  FIG. 2 . The resulting rectified I and Q voltages  12 A and  12 B, respectively, are applied to the MIXER/VGA  10  and are used for biasing the I-Mixer  14  and the Q-Mixer  16 , respectively. More specifically, and referring to the I-Mixer  14 , the rectified I voltage  12 A is applied as a biasing voltage to the opamp  12 , which also receives the input IP and IM signals  3 A from the filter  3 . The opamp  12  receives the biasing voltage  12 A as a common-mode target voltage (V cm ) and adjusts its output so that the FB cm  (common-mode feedback) input voltage is equal to the biasing voltage. The result is that the current consumption of the I-mixer  14  is given by the expression: (Biasing_Voltage*2)/R 1 . The current difference for the mixer Gilbert cell  14 A (transistors T 5 –T 8 ) is, however, controlled by the differential input signal IP-IM output from the opamp  12 , via resistors R 4 A–R 4 D. It can thus be appreciated that the change in the bias does not alter the output of the I-Mixer  18 . As the value of the input I channel signal decreases the magnitude of the biasing voltage  12 A also decreases, which in turn reduces Vcm and thus reduces the current consumption of the I-Mixer  14 , which is the desired result. 
   Such common-mode feedback (CMFB) techniques can be used to stabilize common-mode voltages for fully differential analog systems by way of adjusting the common-mode output currents. A fully differential amplifier, as in the case of the opamp  12 , offers the opportunity to control the output common-mode voltage independently of the differential voltage. The purpose of the V cm  input is to set the output common-mode voltage. In a typical case the output common-mode voltage can be set at a value that is mid-rail, i.e., mid-way between the two power supplies of the opamp. In general, the voltage difference between the (+) and (−) inputs of the opamp  12  is the input differential voltage V ID . The average of the two input voltages is the input common-mode voltage V IC . The difference between the voltages at the (+) and (−) outputs of the opamp  12  is the output differential voltage V OD . The output common-mode voltage V OC , expressed as FBcm in the case of the opamp  12 , is the average of the two voltages at the (+) and (−) outputs of the opamp  12 , and is controlled by the voltage (biasing voltage  12 A) applied to the V cm  input. The above-mentioned opamp feedback path can be seen to be established to the opamp FB cm  input via resistors R 2 A and R 2 B, which are tied between the emitters of the Gilbert cell transistors T 5 , T 6  and T 7 , T 8 . 
   As was noted above, the FB cm  input node of the opamp  12  is used in conjunction with the V cm  input node. The opamp  12  operates so as to change the common mode voltage of the outputs OUTP and OUTM (of the Gilbert cell  14 A) such that the FB cm  node voltage equals the V cm  target (biasing) voltage. Approximate resistor values may be, as examples only, R 1 &lt;100 Ohms, R 2 &gt;20 kOhms, R 3 =Rinp=Rinm&lt;20 kOhms, R 4 &gt;1 kOhms, assuming opamp  12  supply voltages in the range of, for example, zero volts to five volts (or less). The voltage difference (V diff ) between the top nodes of resistors R 1 A and R 1 B (i.e., the nodes that are also connected to the emitters of T 5 ,T 6  and T 7 ,T 8 , and to feedback resistors R 2 A and R 2 B) can be represented by V diff =(R 3 /RIN)*VIN. If the resistors are valued such that R 3 A=R 3 B=Rinp=Rinm, then the voltage difference between IM and IP is transferred to the emitters of T 5 ,T 6  and T 7 ,T 8 . 
   The current of the MIXER/VGA  10  can be significantly reduced through the use of this invention. It should be noted, however, that this invention can be used to reduce the current consumption in other circuit blocks as well, such as in the PA  6 , so long as the block is not sensitive to amplitude modulation. 
   In accordance with this invention, the current consumption of the MIXER/VGA  10  is controlled with the input signal (the I or Q signal, depending on whether one is considering the I-Mixer  14  or the Q-Mixer  16 ). The resulting reduction in current can exceed, for example, 30%. This change in bias has not been possible to achieve with the conventional transmitter topologies, such as the one shown in  FIG. 1 , since the change in bias results in amplitude modulation of the transmitted signal. However, by using the single stage MIXER/VGA  10  the biasing is accomplished using the input opamp  12  and, therefore, all of the undesirable parasitic effects that might arise from the change in bias are divided by the gain of the opamp  12 . The gain of the transmitter chain  1 A can thus be made linear, and current consumption can be beneficially reduced when operating with lower input signals. 
   The foregoing embodiments should be viewed as being exemplary of this invention, and not as imposing limitations upon the practice of this invention. For example, the various component values, component interconnections, signal bandwidths, frequencies and the like are all provided as examples, and are not to be read or construed in a limiting sense. Also by example, while the invention has been described in the context of circuitry having differential inputs and outputs, those skilled in the art should appreciate that at least certain aspects of this invention can be applied as well to non-differential, single ended circuitry. Furthermore, and as was noted, the use of this invention is not limited to reducing current consumption in only the transmitter mixer/VGA. 
   Thus, while these teachings have been particularly shown and described with respect to preferred embodiments thereof, it will be understood by those skilled in the art that changes in form and details may be made therein without departing from the scope and spirit of these teachings.