Abstract:
A channel estimator for determining channel weighting coefficients for a finger of the RAKE receiver. The channel estimator comprises: 1) a first correlator for receiving a first pilot channel signal and correlating the first pilot channel signal with a first pilot channel chip pattern associated with the first pilot channel signal to produce an output comprising a first pilot channel symbol sequence; and 2) a first channel estimation filter capable of receiving the first pilot channel symbol sequence and generating first channel weighting coefficients. The first channel estimation filter minimizes the mean squared error of the channel estimate in the first channel weighting coefficients caused by additive noise and Doppler effects, wherein the mean squared error is minimized across a range of Doppler frequencies from 0 Hz up to a predetermined maximum Doppler frequency.

Description:
TECHNICAL FIELD OF THE INVENTION 
   The present invention relates generally to wireless receivers and, more particularly, to an apparatus and a related method in a wireless receiver that performs channel estimation using minimum mean squared error (MMSE) across a range of Doppler frequencies. 
   BACKGROUND OF THE INVENTION 
   Business and consumers use a wide array of wireless devices, including cell phones, wireless local area network (LAN) cards, global positioning system (GPS) devices, electronic organizers equipped with wireless modems, and the like. The increased demand for wireless communication devices has created a corresponding demand for technical improvements to such devices. Generally speaking, wireless system designers attempt to minimize the cost of conventional radio receivers while improving the performance of such devices. Performance improvements include, among other things, lower power consumption, greater range, increased receiver sensitivity, lower bit error rates (BER), higher transmission rates, and the like. 
   Signal fading due to variations in channel characteristics is a major factor limiting the performance of modern mobile wireless communication systems. To compensate for signal fading, many modern code division multiple access (CDMA) networks use diversity techniques to transmit multiples copies of a signal over a channel to a mobile station. In the mobile station, a RAKE receiver uses multiple baseband correlators to individually process several signal multipath components. The correlator outputs are then combined to achieve improved performance. 
   However, a RAKE receiver assumes that the channel variations over time are relatively slow. This may not be the case if the mobile station is moving relatively quickly. Channel variations due to Doppler effects caused by the relative motion of the base station transmitter and the mobile station receiver often become significant. To correct this, receivers have been developed that use minimum mean squared error (MMSE) channel estimation filters that require knowledge of the specific Doppler frequency and the specific signal-to-interference ratio (SIR) level. Unfortunately, if the Doppler frequency or the SIR level changes, the filter also changes. Thus, the RAKE receiver requires a Doppler estimator and each finger of the RAKE receiver requires its own SIR estimator. 
   Therefore, there is a need in the art for improved radio frequency (RF) receivers. In particular, there is a need for improved channel estimation filters for use in RAKE receivers. More particularly, there is a need for a MMSE channel estimation filter that is not specific to a particular Doppler frequency or SIR level provided the SIR is maintained within a reasonable limit by use of downlink power control. 
   SUMMARY OF THE INVENTION 
   The present invention comprises a low complexity channel estimation filter for a DS-CDMA RAKE receiver that is optimized to work on a range of Doppler frequencies using the average MMSE criterion. The filter structure of the present invention remains the same at all Doppler frequencies and SIR levels and is chosen to perform best on the ensemble average of all Doppler frequencies. 
   Channel estimation is performed on each RAKE finger in order to compensate for the complex channel gain that is associated with each multipath in a scattering environment. In order to estimate the channel gain, a known signal is required. The common pilot channel (CPICH) is the phase reference for all common channels and dedicated channels transmitted throughout the cell. The S-CPICH is the phase reference for dedicated channels transmitted using beam forming. The present invention uses whichever CPICH is the phase reference to estimate the channel gain. 
   Multiplying the de-spread data symbols by the conjugate of the channel gain can perform the channel compensation as well as weighting the inputs for maximal ratio combining. Channel estimation involves finding the mean of a non-stationary time series of de-spread CPICH symbols. The present invention uses a filter that effectively reduces the error in the estimate due to additive noise, while having a low delay for following the mean. In an advantageous embodiment, the present invention may be implemented entirely using a digital signal processor (DSP). 
   The method for the channel estimation follows the following steps: 
   1) Determine the ideal filter characteristics of a filter that minimize the mean square error in the channel estimate. This is done by assuming the Jake&#39;s spectrum for a particular Doppler frequency, taking the channel estimation mean square error spectrum at that Doppler, taking the mean of the mean squared error spectrum over all Doppler frequencies, and minimizing the result over the filter transfer function. 
   2) A discrete-time approximation of the obtained transfer function is performed using the MMSE criterion in order to derive a realizable filter at low complexity. According to an advantageous embodiment of the present invention, such a filter may be implemented as a cascade of a 3-tap finite impulse response (FIR) filter and a single pole infinite impulse response (IIR) filter. Such an embodiment is particularly useful because the operations involved can be performed in one cycle of any DSP with 4 simultaneous MAC operations. 
   3) A fixed-point realization of the filter was obtained where the filter coefficients as well as the data were 16-bit quantized. The quantization levels are chosen to minimize the overall degradation in the signal-to-noise ration (SNR) of the signal in overflow and underflow conditions. 
   To address the above-discussed deficiencies of the prior art, it is a primary object of the present invention to provide, for use in a RAKE receiver, a channel estimator for determining channel weighting coefficients for a finger of the RAKE receiver. According to an advantageous embodiment, the channel estimator comprises: 1) a first correlator for receiving a first pilot channel signal and correlating the first pilot channel signal with a first pilot channel chip pattern associated with the first pilot channel signal to produce an output comprising a first pilot channel symbol sequence; and 2) a first channel estimation filter capable of receiving the first pilot channel symbol sequence and generating first channel weighting coefficients, wherein the first channel estimation filter minimizes a mean squared error in the first channel weighting coefficients caused by additive noise and variation in the channel (Doppler effects), wherein the mean squared error is minimized across a range of Doppler frequencies from 0 Hz up to a predetermined maximum Doppler frequency. 
   According to one embodiment of the present invention, the first channel estimation filter comprises a cascade of a 3-tap finite impulse response (FIR) filter and a single pole infinite impulse response (IIR) filter. 
   According to another embodiment of the present invention, the single pole infinite impulse response (IIR) filter is a Butterworth filter having a maximally flat passband. 
   According to still another embodiment of the present invention, the channel estimator further comprises: 3) a second correlator for receiving a second pilot channel signal and correlating the second pilot channel signal with a second pilot channel chip pattern associated with the second pilot channel signal to produce an output comprising a second pilot channel symbol sequence; and 4) a second channel estimation filter capable of receiving the second pilot channel symbol sequence and generating second channel weighting coefficients, wherein the second channel estimation filter minimizes a mean squared error in the second channel weighting coefficients caused by additive noise and Doppler effects, wherein the mean squared error is minimized across a range of Doppler frequencies from 0 Hz up to a predetermined maximum Doppler frequency. 
   According to a further embodiment of the present invention, the second channel estimation filter comprises a cascade of a 3-tap finite impulse response (FIR) filter and a single pole infinite impulse response (IIR) filter. 
   According to a still further embodiment of the present invention, the channel estimator as set forth in claim  5  wherein the single pole infinite impulse response (IIR) filter is a Butterworth filter having a maximally flat passband. 
   Before undertaking the DETAILED DESCRIPTION OF THE INVENTION below, it may be advantageous to set forth definitions of certain words and phrases used throughout this patent document: the terms “include” and “comprise,” as well as derivatives thereof, mean inclusion without limitation; the term “or,” is inclusive, meaning and/or; the phrases “associated with” and “associated therewith,” as well as derivatives thereof, may mean to include, be included within, interconnect with, contain, be contained within, connect to or with, couple to or with, be communicable with, cooperate with, interleave, juxtapose, be proximate to, be bound to or with, have, have a property of, or the like; and the term “controller” means any device, system or part thereof that controls at least one operation, such a device may be implemented in hardware, firmware or software, or some combination of at least two of the same. It should be noted that the functionality associated with any particular controller may be centralized or distributed, whether locally or remotely. Definitions for certain words and phrases are provided throughout this patent document, those of ordinary skill in the art should understand that in many, if not most instances, such definitions apply to prior, as well as future uses of such defined words and phrases. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention and its advantages, reference is now made to the following description taken in conjunction with the accompanying drawings, in which like reference numerals represent like parts: 
       FIG. 1  illustrates an exemplary wireless network in which mobile station RAKE receivers using channel estimation techniques according to the principles of the invention may be used. 
       FIG. 2  is a timing diagram illustrating the modulation pattern for the common pilot channel (CPICH) signals in the wireless network in  FIG. 1  according to an exemplary embodiment of the present invention; 
       FIG. 3  is a high-level block diagram of a RAKE receiver in an exemplary mobile station according to one embodiment of the present invention; 
       FIG. 4  is a block diagram of a channel estimator according to an exemplary embodiment of the present invention; 
       FIG. 5  is a graph illustrating the power spectral density of the received signal averaged over the entire range of Doppler frequencies when the vehicle speed is a uniformly distributed random variable (with maximum Doppler frequency of 500 Hz) and the ideal channel estimation filter, G(f), in the wireless network; 
       FIG. 6  illustrates a channel estimation filter for calculating weighting coefficients in a RAKE receiver according to an exemplary embodiment of the present invention; 
       FIG. 7  illustrates a pole-zero plot for a channel estimation filter in accordance with the principles of the present invention; 
       FIG. 8  illustrates a symbol de-rotator according to an exemplary embodiment of the present invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIGS. 1 through 8 , discussed below, and the various embodiments used to describe the principles of the present invention in this patent document are by way of illustration only and should not be construed in any way to limit the scope of the invention. Those skilled in the art will understand that the principles of the present invention may be implemented in any suitably arranged mobile station RAKE receiver. 
     FIG. 1  illustrates exemplary wireless network  100 , in which mobile station RAKE receivers using channel estimation techniques according to the principles of the present invention may be used. Wireless network  100  comprises a plurality of cell sites  121 - 123 , each containing a base station (BS), such as BS  101 , BS  102 , or BS  103 . Base stations  101 - 103  communicate with a plurality of mobile stations (MS)  111 - 114  over, for example, code division multiple access (CDMA) channels. Mobile stations  111 - 114  may be any suitable wireless devices, including conventional cellular radiotelephones, Personal Communication Services (PCS) handset devices, personal digital assistants, portable computers, or metering devices. The present invention is not limited to mobile devices. Other types of access terminals, including fixed wireless terminals, may be used. However, for the sake of simplicity, only mobile stations are shown and discussed hereafter. 
   Dotted lines show the approximate boundaries of the cell sites  121 - 123  in which base stations  101 - 103  are located. The cell sites are shown approximately circular for the purposes of illustration and explanation only. It should be clearly understood that the cell sites may have other irregular shapes, depending on the cell configuration selected and natural and man-made obstructions. 
   As is well known in the art, cell sites  121 - 123  are comprised of a plurality of sectors (not shown), each sector being illuminated by a directional antenna coupled to the base station. The embodiment of  FIG. 1  illustrates the base station in the center of the cell. Alternate embodiments position the directional antennas in corners of the sectors. The system of the present invention is not limited to any particular cell site configuration. 
   In one embodiment of the present invention, BS  101 , BS  102 , and BS  103  comprise a base station controller (BSC) and one or more base transceiver subsystem(s) (BTS). Base station controllers and base transceiver subsystems are well known to those skilled in the art. A base station controller is a device that manages wireless communications resources, including the base transceiver stations, for specified cells within a wireless communications network. A base transceiver subsystem comprises the RF transceivers, antennas, and other electrical equipment located in each cell site. 
   BS  101 , BS  102  and BS  103  transfer voice and data signals between each other and the public switched telephone network (PSTN) (not shown) and the Internet via communication line  131 , mobile switching center (MSC)  140 , and packet data serving node (PDSN)  150 . MSC  140  is a switching device that provides services and coordination between the subscribers in a wireless network and external networks, such as the PSTN or Internet. 
   In the exemplary wireless network  100 , MS  111  is located in cell site  121  and is in communication with BS  101 . MS  113  is located in cell site  122  and is in communication with BS  102 . MS  114  is located in cell site  123  and is in communication with BS  103 . MS  112  is also located close to the edge of cell site  123  and is moving in the direction of cell site  123 , as indicated by the direction arrow proximate MS  112 . At some point, as MS  112  moves into cell site  123  and out of cell site  121 , a hand-off will occur. 
   The base stations may transmit from a single antenna or from two antennas. If two antennas are used, the base stations may use transmit diversity (e.g., space-time transmit diversity (STTD)) by coding data in a space-time code and transmitting the pilot symbols in an orthogonal pattern, such as the pattern illustrated in  FIG. 2 . 
     FIG. 2  illustrates timing diagram  200 , which depicts the modulation pattern for the common pilot channel (CPICH) signals in wireless network  100  according to an exemplary embodiment of the present invention. In  FIG. 1 , each of BS  101 -BS  103  has two antennas that may be used to communicate with MS  111 -MS  114 . Each of base stations  101 - 103  may use a single antenna to communicate in a non-transmission diversity (non-TD) mode with the mobile stations. However, in an advantageous embodiment of the present invention, each of base stations  101 - 103  may combat the effects of multipath fading by transmitting from two antennas in a space-time transmit diversity (STTD) mode. 
   In an exemplary embodiment, wireless network  100  is compatible with the 3 rd  Generation Partnership Project (3GPP) standard. In a 3GPP system, during non-TD mode, a common pilot channel (CPICH) signal is transmitted as a quadrature signal from a single antenna using the pattern shown for Antenna  1  in  FIG. 2 , where A=1+j. During STTD mode, a first common pilot channel (CPICH) signal is transmitted as a first quadrature signal from a first antenna using the pattern shown for Antenna  1  in  FIG. 2 , and a second common pilot channel (CPICH) signal is transmitted as a second quadrature signal from a second antenna using the pattern shown for Antenna  2  in  FIG. 2 . 
     FIG. 3  is a high-level block diagram of RAKE receiver  300  in exemplary mobile station  111  according to one embodiment of the present invention. RAKE receiver comprises antenna  301 , radio frequency (RF) front-end block  305 , L fingers, including exemplary fingers  310 ,  320  and  330 , and combiner  340 . Finger  310  comprises delay element  311 , multiplier  312 , summer  313  and multiplier  314 . Finger  320  comprises delay element  321 , multiplier  322 , summer  323  and multiplier  324 . Finger  330  comprises delay element  331 , multiplier  332 , summer  333  and multiplier  334 . 
   RF front-end block  305  downconverts the incoming RF signals received from antenna  301  and produces a baseband or intermediate frequency signal, which is sampled and quantized by an analog-to-digital converter (ADC) to produce a sequence of digital values, the signal R. The signal R is supplied as the input to each of the L fingers. In each of the L fingers, there is a correlator formed by a multiplier and a summer. For example, in finger  310 , the correlator is formed by multiplier  312  and summer  313 , in finger  320 , the correlator is formed by multiplier  322  and summer  323 , and in finger  330 , the correlator is formed by multiplier  332  and summer  333 . 
   In each finger, the signal R is initially delayed by some time delay D(n) by the delay elements. The output of each delay element is the input of the correlator for that finger. Thus, the correlators are synchronized to each of the L strongest multipath components by delaying the received signal R in each finger by an appropriate amount of time D(n). The delayed samples of the received signal R are then correlated with the chip pattern, c(k), to produce a correlated output. The correlated outputs of the correlators are then weighted by coefficients b(n) by the multipliers  314 ,  324 , and  334 . Combiner  340  combines the weighted outputs  340  and the resulting DATA OUT signal is the baseband signal. 
   According to the principles of the present invention, the weighting coefficients b(n) in each of the L fingers of RAKE receiver  300  are calculated by a channel estimation filter that uses the pilot channel signals transmitted by base stations  101 ,  102 , and  103  and that optimizes the weighting coefficients b(n) over a range of Doppler frequencies using the average MMSE criterion. In an advantageous embodiment of the present invention, a digital signal processor (DSP) performs channel estimation. 
     FIG. 4  is a block diagram of channel estimator  400  according to an exemplary embodiment of the present invention. There is one channel estimator for each active finger. The inputs to channel estimator  400  are the integrated CPICH symbols and the CPICH symbol pattern(s) for each antenna. The output(s) are the channel estimates for each antenna-to-antenna path. 
   Channel estimator  400  comprises multipliers  410 ,  420 A and  420 B, integrate and dump blocks  430 A and  430 B and channel estimation filters  440 A and  440 B. It is noted that the elements in  FIG. 4  may be actual circuits in a fixed embodiment. However, if the RAKE receiver is implemented in a digital signal processor (DSP), the elements in  FIG. 4  may be logical functional blocks, rather than literal circuits. 
   Multiplier  410  receives pilot channel signal(s), CPICH, from a pilot signal correlator. In non-TD mode, a single pilot channel signal is received as a sequence of complex number symbols. In STTD mode, two pilot channel signals are received as a sequence of complex number symbols. Multiplier  410  multiplies the CPICH input by the complex number (1+j)/2. The output of multipliers  410  is sent to two correlators. The first correlator comprises multiplier  420 A, integrate and dump block  430 A, and channel estimation filter  440 A. The first correlator produces the channel estimates, H 1 , a complex number. The second correlator comprises multiplier  420 B, integrate and dump block  430 B, and filter  440 B. The second correlator produces the channel estimates, H 2 , a complex number. H 1  and H 2  are used as the weighting coefficients (i.e., b(n)) for the fingers of the RAKE receiver. 
   Multiplier  420 A multiplies the complex symbol output of multiplier  410  by the pilot channel chip pattern for Antenna  1  (i.e., ANT.  1  CPICH PATTERN). The output of multiplier  420 A is input to integrate and dump block  430 A, which integrates two symbols at a time (i.e., 256 chips per symbol) and outputs (dumps) the integrated symbol pairs (i.e., 512 chips) to channel estimation filter  440 A. 
   In STTD mode, multiplier  420 B multiplies the complex symbol output of multiplier  410  by the pilot channel chip pattern for Antenna  2  (i.e., ANT.  2  CPICH PATTERN). The output of multiplier  420 B is input to integrate and dump block  430 B, which integrates two symbols at a time (i.e., 256 chips per symbol) and outputs (dumps) the integrated symbol pairs (i.e., 512 chips) to channel estimation filter  440 B. 
   Channel estimation is required to be performed on each RAKE finger in order to compensate for the complex channel gain associated with each multipath in a scattering environment. In order to estimate the channel gain, a known signal is required. The common pilot channel P-CPICH is the phase reference for all common channels and dedicated channels transmitted throughout the cells in wireless network  100 . The common pilot channel S-CPICH is the phase reference for dedicated channels transmitted using beam forming. In either event, whichever CPICH is present is used as the phase reference for estimating the channel gain. 
   Multiplying the despreaded data symbols by the conjugate of the channel gain can perform the channel compensation as well as weighting the inputs for maximal ratio combining. Fundamentally, the process of channel estimation is an estimation problem of finding the mean of a nonstationary time series of despreaded CPICH symbols. The objective is to use a filter that effectively reduces the error in the estimate due to additive noise, while having a low delay for following the mean value. 
   The starting point for constructing such a channel estimation filter is to consider the ideal characteristics expected from a channel estimation filter. The combination of the channel gains on each of the RAKE fingers define the entire channel (almost completely) as: 
   
     
       
         
           
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   After passing through this channel, the received and integrated pilot channel symbols can be represented as:
 
 r   l ( t )= A   p (1+ j )δ( n−d   l )*( h ( n,k )+η( n,k )) l  
 
   The channel estimator derotates the original 45° rotation in the pilot symbols and filters the output with a real filter. This operation may be represent by: 
   
     
       
         
           
             
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   The channel gain estimators act on the pilot signals demodulated by each RAKE finger. The pilot signals are perturbed by noise. Thus, the output of each channel estimator is:
 
( ĉ   r ( k )+ jĉ   i ( k )) l   =A   p   g ( k )*δ( n−d   l )*( h ( n,k )+η( n,k ))
 
           ( ĉ   r ( k )+ jĉ   i ( k )) l   =A   p   g ( k )*( c   r ( k )+ jc   i ( k )+η( k )) l  
 
where A p  is the pilot amplitude, r(n) is the rake finger output and g(n) is the channel estimator.

   The noise process perturbing the real and imaginary component of the channel gain on each finger is Gaussian (or nearly so). Hence, the optimum criterion for the channel gain estimators on each finger is to minimize the mean square error (MSE) of the estimates.
 
 e   l ( k )=[( c   r ( k )+ jc   i ( k )) l −( ĉ   r ( k )+ jĉ   i ( k )) l ]
 
   By Parseval&#39;s theorem, minimizing MSE in the time domain is equivalent to minimizing MSE in the frequency domain. In the frequency domain, the following model may be considered:
 
 Ĥ ( f )= G ( f )[ H ( f )+ N ( f )]
 
   The problem is to find G(f) such that: 
                 G   ⁡     (   f   )       ⁢     :     ⁢           ⁢       min     G   ⁡     (   f   )         ⁢       ∫       f   m     =   0       f   max       ⁢       p   ⁡     (     f   m     )       ⁢     E   θ     ⁢     {              H   ⁡     (     f   ,     f   m     ,   θ     )       -       H   ^     ⁡     (   f   )              2     }     ⁢           ⁢     ⅆ     f   m               =       min     G   ⁡     (   f   )         ⁢           ⁢       E     f   m       ⁢     {       E   θ     ⁢     {              H   ⁡     (     f   ,     f   m     ,   θ     )       -       H   ^     ⁡     (   f   )              2     }       }           ,         
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   In order to proceed further, it is necessary to assume a probability density function (PDF) for Doppler frequency. It may be assumed (for lack of any definitive data) that the velocity of the mobile stations may be distributed uniformly from 0 kilometers per hour (kmph) to 250 kmph. Then: 
   
     
       
         
           
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   This is plotted as in  FIG. 5 .  FIG. 5  illustrates graph  500 , which depicts the power spectral density of the received signal averaged in curve  505  over the entire range of Doppler frequencies when the vehicle speed is an uniformly distributed random variable (with maximum Doppler frequency of 500 Hz) and the ideal channel estimation filter, G(f), in curve  510 . As can be seen, the fall is quite sharp at 500 Hz. In fact, there are no components beyond 500 Hz, which roughly corresponds to the maximum Doppler frequency at 2 GHz when the mobile speed is 250 kmph. 
   For a non-TD system, the sampling frequency is 15 kHz, corresponding to integration of 256 pilot chips. Hence, a digital filter needs to be constructed that closely approximates the G(f) in  FIG. 5  with a Nyquist frequency of 7.5 kHz, thereby satisfying complexity constraints. 
   According to an advantageous embodiment of the present invention, a second-order filter may be used. Since the overriding aim is to have small attenuation in the stopband, an equiripple pass band filter may be used. To keep the stopband attenuation minimum, an equiripple 30 dB stopband and a 1 dB passband attenuation elliptic filter may be used. 
     FIG. 6  illustrates channel estimation filter  440  according to an exemplary embodiment of the present invention. A preferred digital signal processor (DSP) can perform 4 MAC operations simultaneously. A Butterworth filter involves 2 MAC operations, while a elliptic (or any other 2 nd  order) filter involves 5 AMC operations. One effective way to use the DSP to its fullest advantage is to use a 3-tap finite impulse response (FIR) filter followed by a Butterworth stage as shown in  FIG. 6 . 
   Channel estimation filter  440  comprises a FIR filter stage and a Butterworth filter stage (also known as maximally flat low pass filter stage). The filter stage receives the input X(k). The FIR filter stage comprises delay elements  610 A and  610 B, multipliers  620 A,  620 B and  620 C, and adders  630 A and  630 B. The Butterworth Filter stage comprises delay elements  650  and multiplier  660 . Adder  640  adds the output of the FIR filter stage and the Butterworth filter stage to produce the filtered output Y(k). 
   A first order approximation of the filter above will be a Butterworth filter. The 3 dB cutoff frequency of the filter may be chosen to match that of the ideal filter. This is around 300 Hz. 
   The corresponding filter has the form: 
   
     
       
         
           
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   A first order filter will involve two multiplications and additions. A first order filter may approximate the optimal filter based on expected channel psd well, however, it produces a significant attenuation (˜6 dB @ 500 Hz) in the passband when the highest Doppler is in use. 
   However, a cascade of a FIR filter and a Butterworth filter overcomes the problems of a first order Butterworth filter by itself. In the cascade design, the Butterworth filter design may be kept identical to the one described above as it follows the passband quite closely. The FIR stage may be designed to add additional attenuation in the stopband while keeping very low attenuation in the passband. 
   The equal tap FIR stage of:
 
 G   F ( z )=0.333(1+ z   −1   +z   −2 )
 
gives an attenuation of less than 0.13 dB at 500 Hz and more than 10 dB from 3800 Hz and beyond.
 
   Delaying the despreaded data by one sample prior to channel compensation compensates for the extra sample of group delay caused by the FIR stage. Thus, the combined filter is: 
   
     
       
         
           
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               ) 
             
           
           = 
           
             
               
                 0.0395 
                 ⁢ 
                 
                   ( 
                   
                     z 
                     + 
                     1 
                     + 
                     
                       z 
                       
                         - 
                         1 
                       
                     
                   
                   ) 
                 
               
               
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                   0.8816 
                   ⁢ 
                   
                     z 
                     
                       - 
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             . 
           
         
       
     
   
   STTD Mode of Operation 
   The mobile station conducts a secondary search to determine whether the base station is using the space-time transmit diversity (STTD) mode of operation. Once it is determined that STTD mode is being used, a STTD channel estimation technique is used. 
   In the STTD mode there are two channels to be estimated, the channel from the first base station (BS) antenna (i.e., Antenna  1 ) to the mobile station (MS), h 1 (t), and the channel from the second BS antenna to the MS, h 2 (t). In STTD mode of operation the two antennas transmit the pilot channel bit patterns in phase and in anti-phase alternatively, as shown in  FIG. 2 . Hence, the received and integrated CPICH symbols on each finger will follow the pattern:
 
 r   l ( t )= A   p (1+ j )δ( n−d   l )*( h   l ( n,k )+ a   n   h   2 ( n,k )+η( n,k )) l ,
 
where αa n ε{−1,1} forms the phase sequence of the second antenna&#39;s CPICH transmission.
 
   Hence, at any given time the effective received channel is the sum or the difference of the two channels. 
   There are two methods for performing channel estimation: 
   1) Method 1: Estimate the sum and differences of the channel separately by integrating separately over intervals when the CPICH transmissions are in-phase and out-of-phase. The individual channels can then be calculated by taking the sum and the difference; and 
   2) Method 2: Integrate over intervals where the in-phase and out-of-phase intervals are matched equally to produce super-symbols. By choosing the sign of the despreading code, either the first or the second channel may be estimated. 
   The advantages and disadvantages are of these methods are: 
   1) Method 1: The order of pilot symbols where the CPICH is from the first and second antennas are the same (or different) signs is not periodic. This means the first method requires housekeeping for symbol counts. Also, because of this aperiodic characteristic, the filter either must run at symbol rate or have special operations at symbol boundaries. 
   2) Method 2: The sampling frequency gets halved leading to a 50% MIPS or power savings. Also, no housekeeping beyond counting even and odd symbols is necessary. 
   The second approach is preferred in this algorithm. 
   The operation involved is: 
               (           c   ^       1   ⁢   r       ⁡     (   k   )       +     j   ⁢           ⁢         c   ^       1   ⁢   i       ⁡     (   k   )           )     l     =     {             g   ⁡     (   k   )       *     1   2     ⁢       ∑     m   =     n   -   1       n     ⁢           ⁢         (     1   -   j     )     2     ⁢     δ   ⁡     (     m   -     d   l       )       *     r   ⁡     (     m   ,   k     )                   n   =       2   ⁢     ⌊     n   /   2     ⌋       +   1                   g   ⁡     (   k   )       *     1   2     ⁢       ∑     m   =     n   -   2         n   -   1       ⁢           ⁢         (     1   -   j     )     2     ⁢     δ   ⁡     (     m   -     d   l       )       *     r   ⁡     (     m   ,   k     )                   n   =     2   ⁢     ⌊     n   /   2     ⌋                       
for Antenna  1 , and
 
               (           c   ^       2   ⁢   r       ⁡     (   k   )       +     j   ⁢           ⁢         c   ^       2   ⁢   i       ⁡     (   k   )           )     l     =     {             g   ⁡     (   k   )       *     1   2     ⁢       ∑     m   =     n   -   1       n     ⁢           ⁢         (     1   -   j     )     2     ⁢     a   m     ⁢     δ   ⁡     (     m   -     d   l       )       *     r   ⁡     (     m   ,   k     )                   n   =       2   ⁢     ⌊     n   /   2     ⌋       +   1                   g   ⁡     (   k   )       *     1   2     ⁢       ∑     m   =     n   -   2         n   -   1       ⁢           ⁢         (     1   -   j     )     2     ⁢     a   m     ⁢     δ   ⁡     (     m   -     d   l       )       *     r   ⁡     (     m   ,   k     )                   n   =     2   ⁢     ⌊     n   /   2     ⌋                       
for Antenna  2 .
 
   The resulting output from each channel estimator is:
 
( ĉ   sr ( k )+ jĉ   sl ( k )) l   =A   p   g ( k )*δ( n−d   l )*( h   s ( n,k )+η s ( n,k ))s=1,2
 
           ( ĉ   sr ( k )+ jĉ   sl ( k )) l   =A   p   g ( k )*( c   sr ( k )+ jc   sl ( k )+η s ( k )) l  

   Note that due to averaging, the variance of the noise samples η sl (k) is half of that of the non-TD case η l (k). 
   The filter itself can take the same form as in  FIG. 6 . However, because the filtering is taking place over supersymbols (which are the sum of two consecutive symbols), the sampling frequency is halved to 7.5 kHz. Hence, the filter coefficients must be adjusted accordingly. 
   FIR-Butterworth Combination 
   The Butterworth filter in the non-TD design was designed to have a 3 dB cutoff frequency at 300 Hz. For a sampling frequency of 7500 Hz, the corresponding filter will be: 
   
     
       
         
           
             
               G 
               But 
             
             ⁡ 
             
               ( 
               z 
               ) 
             
           
           = 
           
             
               0.11216 
               ⁢ 
               
                 ( 
                 
                   1 
                   + 
                   
                     z 
                     
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                       1 
                     
                   
                 
                 ) 
               
             
             
               1 
               - 
               
                 0.77568 
                 ⁢ 
                 
                   z 
                   
                     - 
                     1 
                   
                 
               
             
           
         
       
     
   
   Note that both the pole position is closer to 0.5, leading to a filter design with lesser vulnerability to quantization error. The combined filter is obtained by cascading the FIR filter and the Butterworth filter stages: 
   
     
       
         
           
             G 
             ⁡ 
             
               ( 
               z 
               ) 
             
           
           = 
           
             0.12818 
             ⁢ 
             
               
                 
                   0.5 
                   + 
                   
                     0.75 
                     ⁢ 
                     
                       z 
                       
                         - 
                         1 
                       
                     
                   
                   + 
                   
                     0.5 
                     ⁢ 
                     
                       z 
                       
                         - 
                         2 
                       
                     
                   
                 
                 
                   1 
                   - 
                   
                     0.77568 
                     ⁢ 
                     
                       z 
                       
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               . 
             
           
         
       
     
   
     FIG. 7  illustrates pole-zero plot  700  for a channel estimation filter in accordance with the principles of the present invention. Note that the pole position is well inside the unit circle leading to a filter design with more stability, better response time, lesser group delay and lesser vulnerability to quantization error. 
   Symbol De-Rotation in STTD Mode 
   In STTD mode, the transmitted symbols are transmitted from Antennas  1  and  2  of the base stations in the following fashion: 
   
     
       
             
             
             
           
         
             
                 
             
             
               Transmission Time 
               Antenna 1 symbol 
               Antenna 2 Symbol 
             
             
                 
             
           
           
             
               2m 
               S 2m   
               −S* 2m+1   
             
             
               2m + 1 
               S 2m+1   
               S* 2m   
             
             
                 
             
           
        
       
     
   
   It is assumed that the first symbol interval in the frame was symbol interval  0 . Hence, the received symbols on even symbol intervals are:
 
 r   2m   =h   1   S   2m   −h   2   S*   2m+1 +η 2m .
 
Similarly, the received symbols on odd symbol intervals are:
 
 r   2m+1   =h   1   S   2m+1   +h   2   S*   2m +η 2m+1 .
 
   The optimal de-rotation for the received symbol on even symbol interval is: 
                     S   ^       2   ⁢   m       =         h   1   *     ⁢     r     2   ⁢   m         +       h   2     ⁢     r       2   ⁢   m     +   1     *                     =         (              h   1          2     +            h   2          2       )     ⁢     S     2   ⁢   m         +     (         h   1   *     ⁢     η     2   ⁢   m         +       h   1     ⁢     η       2   ⁢   m     +   1     *         )                   
The optimal de-rotation on odd symbol interval is:
 
   
     
       
         
           
             
               
                 
                   
                     S 
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                                 ⁢ 
                                 m 
                               
                               + 
                               1 
                             
                           
                         
                       
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                     . 
                   
                 
               
             
           
         
       
     
   
     FIG. 8  illustrates symbol de-rotator  800  according to an exemplary embodiment of the present invention. There is one symbol de-rotator for every physical channel (other than CPICH) in every finger. The inputs to the symbol de-rotator are the integrated traffic channel (common or dedicated) symbols and the channel estimate(s) H 1  and H 2  from channel estimator  440 . The outputs of symbol de-rotator  800  are the channel compensated symbols ready for combining. As in the case of  FIG. 4 , it is noted that the elements in  FIG. 8  may be actual circuits in a fixed embodiment. However, if the RAKE receiver is implemented in a digital signal processor (DSP), the elements in  FIG. 8  may be logical functional blocks, rather than literal circuits. 
   Symbol de-rotator  800  has a Non-TD Output and a STTD Output. In non-TD mode, each channel estimate, H 1 , from channel estimator  400  is complex conjugated by complex conjugate block  810 A and the output is applied to one input of multiplier  815 . The other input of multiplier  815  receives the traffic channel symbols from traffic channel correlator  890 . The output of multiplier  815  is the channel compensated symbols that form the Non-TD Output. 
   In STTD mode, complex conjugate block  810 A and multiplier  815  operate as in non-TD mode. The Non-TD output is applied to a first input of summer  860 . 
   The traffic channel symbols from traffic channel correlator  890  are complex conjugated by complex conjugate block  810 B and the output is applied to serial-to-parallel (S&gt;P) block  820 , which converts the symbol data from serial to parallel. Alternating symbols from S&gt;P block  820  are stored in registers  825  and  830 . Even symbols (i.e., symbol  2   m ) are stored in register  825  and odd symbols are stored in register  830  (i.e., symbol  2   m+ 1). The symbol data in register  830  is transferred to register  840 , but the symbol data in register  825  is negated by inverter  835  and then stored in register  845 . The symbol data in registers  840  and  845  are then read by parallel-to-serial (P&gt;S) block  850 . 
   The serial symbol data from P&gt;S block  850  is multiplied by the channel estimate, H 2 , from channel estimator  440  by multiplier  855 . The channel compensated symbols from multiplier  855  are then combined with the channel compensated symbols from the Non-TD Output to form the STTD Output. 
   The prior art includes a Wiener filter-based MMSE channel estimation that requires knowledge of SIR and Doppler. Hence it requires a Doppler and SIR estimator. The filter structure will change dynamically when these quantities change. The present invention overcomes numerous disadvantages of the prior art including: 
   1) There is no need for a Doppler estimator or a per finger SIR estimator; 
   2) The filter structure does not change with changes in is Doppler and SIR; and 
   3) There is not need for time-intensive calculation, such as matrix inversions. 
   These improvements make the design simpler, consume less power, require less silicon area, and the like. 
   The present invention performs sub-optimally compared to the prior art at a particular Doppler value and SIR setting. However, the present invention gives the best performance for the ensemble average for all Doppler settings and performance simulations demonstrate acceptable performance at the entire range of expected Doppler frequencies. The SIR level is typically maintained constant under closed-loop power control and hence this advantage of the related art is of little practical value. 
   Although the present invention has been described with several embodiments, various changes and modifications may be suggested to one skilled in the art. It is intended that the present invention encompass such changes and modifications as fall within the scope of the appended claims.