Abstract:
An apparatus includes a first estimator that estimates a signal quality based on an error correction number of an electrical signal obtained by photoelectrically converting a received optical signal; a second estimator that estimates a signal quality from which the influence of nonlinear effects is removed based on signals upstream and downstream of an identification calculator identifying the electrical signal; and a calculator that calculates the difference between the signal qualities estimated by the first and second estimators to calculate the amount of nonlinear effects.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
       [0001]    This application is based upon and claims the benefit of priority of the prior Japanese Patent Application No. 2009-245179, filed on Oct. 26, 2009, the entire contents of which are incorporated herein by reference. 
       FIELD 
       [0002]    The embodiments relate to a nonlinear distortion detecting module, an optical receiver, an optical transmission system, and a method for detecting nonlinear distortion. 
       BACKGROUND 
       [0003]    In communication networks, optical communication using optical fibers as transmission path segments is often used. To cope with a recent increase in the amount of information streaming through a network, increasing the transmission distance and transmission capacity of an optical transmission system is required. However, the transmission distance and transmission capacity of the optical transmission system are limited due to optical waveform distortions in optical fibers. The optical waveform distortions include linear distortion and nonlinear distortion. 
         [0004]    The linear distortion includes, for example, chromatic dispersion and polarization mode dispersion. The linear distortion can be compensated for using a digital coherent technique and an equalizing technique based on digital signal processing, or using an optical compensating technique in an optical receiving device receiving an optical signal transmitted through optical fibers. The nonlinear distortion includes, for example, cross-phase modulation and self-phase modulation. Compensation for the nonlinear distortion in an optical receiving device is disclosed in, for example, Kazuro Kikuchi, “Electronic Post-compensation for Nonlinear Phase Fluctuations in a 1000-km 20-Gbit/s Optical Quadrature Phase-shift Keying Transmission System Using the Digital Coherent Receiver”, OPTICS EXPRESS, Vol. 16, No. 2, pp. 889-896, 2008. 
         [0005]    According to the disclosed technique, nonlinear distortion caused in optical fibers can be compensated for in an optical receiving device. However, the above-described technique does not detect nonlinear distortion. Accordingly, the accuracy with which to compensate for nonlinear distortion is limited. 
       SUMMARY 
       [0006]    According to an aspect of the invention, an apparatus includes a first estimator that estimates a signal quality based on an error correction number of an electrical signal obtained by photoelectrically converting a received optical signal; a second estimator that estimates a signal quality from which the influence of nonlinear effects is removed based on signals upstream and downstream of an identification calculator identifying the electrical signal; and a calculator that calculates the difference between the signal qualities estimated by the first and second estimators to calculate the amount of nonlinear effects. 
         [0007]    The object and advantages of the invention will be realized and attained by the elements, features, and combinations particularly pointed out in the claims. 
         [0008]    It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are not restrictive of the invention, as claimed. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0009]      FIG. 1  illustrates an optical transmission system. 
           [0010]      FIG. 2A  illustrates an optical receiver. 
           [0011]      FIG. 2B  illustrates a coherent optical front-end. 
           [0012]      FIG. 3  illustrates an exemplary configuration of an equalizer. 
           [0013]      FIG. 4  illustrates a nonlinear effects monitor. 
           [0014]      FIG. 5  illustrates a control method performed by a controller. 
           [0015]      FIG. 6A  illustrates a nonlinear effects monitor amount detected by a nonlinear effects monitor. 
           [0016]      FIG. 6B  illustrates a result of simulation demonstrating the relationship between the nonlinear effects monitor amount and an actual signal quality degradation amount. 
           [0017]      FIG. 7A  illustrates a change in signal quality. 
           [0018]      FIG. 7B  illustrates a change in the result of detection by the nonlinear effects monitor. 
           [0019]      FIG. 8A  illustrates the spread of a received signal in the vicinity of each signal constellation point. 
           [0020]      FIG. 8B  illustrates a fluctuation X 0 . 
           [0021]      FIG. 9A  illustrates an optical receiver according to a second embodiment. 
           [0022]      FIG. 9B  illustrates a coherent optical front-end. 
           [0023]      FIG. 10A  illustrates an optical receiver according to a third embodiment. 
           [0024]      FIG. 10B  illustrates a coherent optical front-end. 
           [0025]      FIG. 11A  illustrates an optical receiver according to a fourth embodiment. 
           [0026]      FIG. 11B  illustrates a coherent optical front-end. 
       
    
    
     DESCRIPTION OF EMBODIMENTS 
       [0027]    Embodiments will be described below with reference to the drawings. 
       First Embodiment 
       [0028]      FIG. 1  illustrates an optical transmission system  100 . Referring to  FIG. 1 , the optical transmission system  100  includes an optical transmitter  101 , an optical transmission path including a plurality of concatenated transmission spans  102 , and an optical receiver  103 . The optical transmitter  101  outputs an optical signal modulated based on an electrical signal to the optical transmission path. The transmission span  102  at the first stage receives the optical signal from the optical transmitter  101 . This optical signal travels through the transmission spans  102  at multiple stages. The transmission span  102  at the last stage transmits the optical signal to the optical receiver  103 . The optical receiver  103  converts the optical signal to an electrical signal and outputs the electrical signal. 
         [0029]    Each transmission span  102  includes a transmission path segment  104 , an optical amplifier  105 , and a chromatic dispersion compensation module  106 . The transmission path segment  104  is, for example, an optical fiber. The optical amplifier  105  is, for example, a rare earth added optical fiber amplifier or a Raman amplifier and amplifies an optical signal attenuated in the transmission path segment  104 . The chromatic dispersion compensation module  106  compensates for chromatic dispersion caused in the transmission path segment  104 . 
         [0030]      FIGS. 2A and 2B  illustrate the optical receiver  103 . Referring to  FIG. 2A , the optical receiver  103  includes a coherent optical front-end  10 , analog-to-digital converters (ADCs)  20 , a main signal processor  30 , a nonlinear effects monitor  40 , and a controller  50 . 
         [0031]    Referring to  FIG. 2B , the coherent optical front-end  10  is the integration of optical and electronic components and includes a polarization controller  11 , a 90° hybrid  12 , two optical-to-electrical converters (O/Es)  13 , and a local optical oscillator  14 . 
         [0032]    An optical signal received by the optical receiver  103  is input to the polarization controller  11 . An optical signal modulating method applicable to the present embodiment is not particularly limited. The optical signal is, for example, a multi-phase shift keying (mPSK) signal. The polarization controller  11  outputs an optical signal polarized in a desired direction based on the input optical signal. The 90° hybrid  12  demodulates the optical signal based on an oscillation signal from the local optical oscillator  14  and outputs an I-phase signal and a Q-phase signal which are 90 degrees out of phase with each other. 
         [0033]    One of the O/Es  13  converts the I-phase signal to an electrical signal and outputs the resultant signal to one of the ADCs  20 . The other O/E  13  converts the Q-phase signal to an electrical signal and outputs the resultant signal to the other ADC  20 . Each ADC  20  converts the received electrical signal in analog form to a digital electrical signal. The resultant I-phase and Q-phase signals, serving as the digital electrical signals, are input to the main signal processor  30 . 
         [0034]    The main signal processor  30  includes an equalizer  31 , a sub equalizer  32 , a carrier phase recovery circuit  33 , an identification calculator  34 , and an error correction calculator  35 . The equalizer  31  compensates for linear distortion and nonlinear distortion. 
         [0035]      FIG. 3  illustrates an exemplary configuration of the equalizer  31 . Referring to  FIG. 3 , the equalizer  31  includes a plurality of linear distortion compensators  301  and a plurality of nonlinear distortion compensators  302 . One linear distortion compensator  301  is paired with one nonlinear distortion compensator  302 , thus providing one distortion compensator  303 . Either of the linear distortion compensator  301  and the nonlinear distortion compensator  302  may be located at the front stage or back stage of the distortion compensator  303 . The equalizer  31  includes a plurality of distortion compensators  303  connected in series. In the equalizer  31 , therefore, the linear distortion compensators  301  and the nonlinear distortion compensators  302  are alternately arranged. Consequently, the equalizer  31  alternately performs linear distortion compensation and nonlinear distortion compensation on input signals and outputs the distortion-compensated signals. 
         [0036]    The sub equalizer  32  compensates for small distortion that is not compensated for by the equalizer  31  and transmits the distortion-compensated signals to the carrier phase recovery circuit  33 . The carrier phase recovery circuit  33  detects a phase shift between the received signals to recover a signal and transmits the signal to the identification calculator  34 . The identification calculator  34  compares the received signal from the carrier phase recovery circuit  33  with a threshold value in order to identify a signal for each symbol. The error correction calculator  35  compares a redundant portion which can be determined when an error occurs with actual data. If an error occurs, the error correction calculator  35  corrects the error. 
         [0037]    The following is the quotation for explanation of “symbol” from WikipediA on the Internet. Any digital modulation scheme uses a finite number of distinct signals to represent digital data. PSK uses a finite number of phases, each assigned a unique pattern of binary digits. Usually, each phase encodes an equal number of bits. Each pattern of bits forms the symbol that is represented by the particular phase. 
         [0038]    The nonlinear effects monitor  40  detects nonlinear distortion of an optical signal received by the optical receiver  103 . The nonlinear effects monitor  40  will be described in detail later. The controller  50  controls the linear distortion compensators  301  and the nonlinear distortion compensators  302  in the equalizer  31  based on nonlinear distortion detected by the nonlinear effects monitor  40  to compensate for the nonlinear distortion. 
         [0039]    The main signal processor  30  may include, for example, an application specific integrated circuit (ASIC) which is an integrated circuit (IC). This ASIC may include at least either of the ADCs  20 , the nonlinear effects monitor  40 , and the controller  50 . Instead of the ASIC, for example, a field programmable gate array (FPGA) or a digital signal processor (DSP) may be used. Alternatively, the main signal processor  30  may include the ASIC, the FPGA, and the DSP in combination. 
         [0040]    The nonlinear effects monitor  40  will be described in detail below.  FIG. 4  illustrates the nonlinear effects monitor  40 . Referring to  FIG. 4 , the nonlinear effects monitor  40  includes a first estimator  41 , a second estimator  42 , and an adder  43 . The first estimator  41  includes an error-correction-number-to-signal-quality-index converter  401 . The second estimator  42  includes adders  402  and  408 , a squarer  403 , averagers  404  and  407 , a delayer  405 , a multiplier  406 , and a signal quality index calculator  409 . 
         [0041]    The adder  402  receives signals from the carrier phase recovery circuit  33  and the identification calculator  34 . 
         [0042]    The I-phase signal and the Q-phase signal supplied to the carrier phase recovery circuit  33  can be expressed using a scalar (ranging from 0 to 2π) of the phase expressed by the following expression. 
         [0000]      d k +θ k +n k +δ k   (1)
 
         [0043]    Where 
         [0044]    d k : modulated data of the kth symbol (for example, a quadrature value (0° to 270°)); 
         [0045]    θ k : phase noise and offset of the kth symbol; 
         [0046]    n k : white Gaussian noise of the kth symbol; and 
         [0047]    δ k : nonlinear noise of the kth symbol. 
         [0048]    The carrier phase recovery circuit  33  outputs a signal obtained by subtracting a phase angle estimated by signal processing from Expression (1). Specifically, the estimated phase angle is defined as θ k   (est) . 
         [0049]    In addition, θ k −θ k   (est) =Δθ k . In this case, an output signal of the carrier phase recovery circuit  33  can be expressed by the following expression. 
         [0000]      d k +Δθ k +n k +δ k   (2)
 
         [0050]    The identification calculator  34  outputs the modulated data d k . The adder  402  outputs a signal obtained by subtracting an output value of the identification calculator  34  from an output value of the carrier phase recovery circuit  33 . Therefore, the output signal of the adder  402  can be expressed by the following expression. 
         [0000]      Δθ k +n k +δ k   (3)
 
         [0051]    The squarer  403  squares an output value of the adder  402  and outputs the resultant signal to the averager  404 . 
         [0052]    The output signal of the squarer  403  can be expressed by the following expression. 
         [0000]      Δθ k   2 +n k   2 +δ k   2 +2Δθ k n k +2n k δ k +2Δθ k δ k   (4)
 
         [0053]    The averager  404  outputs the average of output signals of the squarer  403  as a signal S 1  to the adder  408 . The white Gaussian noise n k  has no correlation between symbols. Accordingly, when an average for a plurality of symbols is calculated, 2Δθ k n k  and 2n k δ k  in Expression (4) can be regarded as zero. In addition, the nonlinear noise δ k  is random. When an average for a plurality of symbols is calculated, therefore, 2Δθ k δ k  in Expression (4) can also be regarded as zero. 
         [0054]    The delayer  405  delays the output signal of the adder  402  by one symbol and outputs the resultant signal. The multiplier  406  outputs the product of the output signal of the adder  402  and that of the delayer  405  to the averager  407 . An output signal of the multiplier  406  can be expressed by the following expression. 
         [0000]      Δθ k Δθ k+1 +n k n k+1 +δ k δ k+1 +Δθ k n k+1 +Δθ k+1 n k +n k δ k+1 +n k+1 δ k +Δθ k δ k+1 +Δθ k+1 δ k   (5)
 
         [0055]    The averager  407  outputs the average of output signals of the multiplier  406  as a signal S 2  to the adder  408 . The white Gaussian noise n k  has no correlation between symbols as described above. Accordingly, when an average for a plurality of symbols is calculated, n k n k+1 , Δθ k n k+1 , n k δ k+1 , and n k+1 δ k  can be regarded as zero. In addition, the nonlinear noise δ k  is also random. Accordingly, when an average for a plurality of symbols is calculated, Δθ k Δθ k+1  and Δθ k+1 δ k  in Expression (5) can also be regarded as zero so long as it is approximated that Δθ k =Δθ k+1 . The nonlinear noise δ k  is random but has a correlation between symbols. Therefore, if an average for symbols is calculated, δ k δ k+1  in Expression (5) is not zero in some cases. Consequently, an output signal of the adder  408 , which outputs a signal obtained by subtracting the signal S 2  from the signal S 1 , can be expressed by the following expression. 
         [0000]        S 1 −S 2=average( n   k   2 +δ k   2 −δ k δ k+1 )  (6)
 
         [0056]    A bit error rate (BER) is calculated from an intra-frame error correction number. The BER can be converted to a phase angle spread average(n k   2 +δ k   2 ) caused by noise. The phase angle spread caused by noise is subtracted from Expression (6), thus obtaining average(δ k δ k+1 ). Assuming that δ k+1 =δ k +Δ k  and average(Δ k )=0, average(δ k δ k+1 )=average (δ k (δ k +Δ k ))=average(δ k   2 +δ k Δ k )=average (δ k   2 ). Therefore, an amount proportional to the magnitude of nonlinear effects (hereinafter, called “nonlinear effects monitor amount”) can be detected. As described above, a nonlinear effects monitor amount can be separated using the fact that white Gaussian noise has no correlation between symbols and nonlinear noise has a correlation between symbols. 
         [0057]    In the present embodiment, a Q value is used as a signal quality index. Specifically, the error-correction-number-to-signal-quality-index converter  401  converts a BER to a Q value. The Q value can be expressed as √2erfc −1 (2×BER). The signal quality index calculator  409  calculates a signal quality index (Q value) based on the output of the adder  408 . In this case, the Q value can be expressed as Q=(1/2)×1/(S 1 −S 2 ). The adder  43  subtracts the Q value obtained by the error-correction-number-to-signal-quality-index converter  401  from the Q value obtained by the signal quality index calculator  409 . Thus, an amount proportional to the magnitude of nonlinear effects can be detected. 
         [0058]    In this embodiment, a nonlinear effects monitor amount can be acquired using signals upstream and downstream of the identification calculator  34  and an error correction number. Accordingly, a nonlinear effects monitor amount can be acquired using a main signal processor of an existing optical receiver. 
         [0059]    The result of output of the adder  43  is supplied to the controller  50 . The controller  50  controls the equalizer  31  so that the nonlinear effects monitor amount detected by the nonlinear effects monitor  40  decreases. Consequently, nonlinear distortion can be compensated for with high accuracy. In addition, when the controller  50  controls the equalizer  31  so as to minimize the nonlinear effects monitor amount detected by the nonlinear effects monitor  40 , the nonlinear distortion can be maximally compensated for. 
         [0060]    To increase the accuracy, the error-correction-number-to-signal-quality-index converter  401  may perform integration until an error correction number n exceeds a threshold value N. In this case, preferably, the averagers  404  and  407  each output an average up to the extent that the error correction number n exceeds the threshold value N. 
         [0061]      FIG. 5  illustrates a control method performed by the controller  50 . Referring to  FIG. 5 , the controller  50  sets an initial value in the equalizer  31  (operation S 1 ). The controller  50  subsequently acquires a nonlinear effects monitor amount from the nonlinear effects monitor  40  (operation S 2 ). 
         [0062]    Next, the controller  50  estimates the direction of parameter control for decreasing the nonlinear effects monitor amount (operation S 3 ). The controller  50  then controls parameters of the equalizer  31  so that the nonlinear effects monitor amount decreases (operation S 4 ). 
         [0063]    After that, the controller  50  determines whether an output of the nonlinear effects monitor  40  is less than a predetermined value μ (operation S 5 ). If “No” in operation S 5 , the controller  50  again executes the operation S 2  and the subsequent operations. If “Yes” in operation S 5 , the controller  50  terminates the execution of the control method. According to the control method of  FIG. 5 , the nonlinear effects monitor amount detected by the nonlinear effects monitor  40  can be controlled to a small value. Thus, nonlinear distortion can be compensated for. 
         [0064]      FIGS. 6A and 6B  illustrate a result of simulation demonstrating the relationship between the nonlinear effects monitor amount detected by the nonlinear effects monitor  40  and an actual amount of signal quality degradation (hereinafter, “signal quality degradation amount”). Referring to  FIG. 6A , the axis of abscissas denotes an actual nonlinear effects amount, the left axis of ordinates denotes the nonlinear effects monitor amount detected by the nonlinear effects monitor  40 , and the right axis of ordinates denotes the signal quality degradation amount. As illustrated in  FIG. 6A , the nonlinear effects monitor amount and the signal quality degradation amount substantially correspond to each other relative to the actual nonlinear effects amount. In addition, as illustrated in  FIG. 6B , the one-to-one correspondence between the nonlinear effects monitor amount and the signal quality degradation amount is held. 
         [0065]      FIGS. 7A and 7B  illustrate a signal quality obtained when the controller  50  controls the equalizer  31  based on the result of detection by the nonlinear effects monitor  40 .  FIG. 7A  illustrates a change in signal quality.  FIG. 7B  illustrates a change in the result of detection by the nonlinear effects monitor  40 . The axis of abscissas in each of  FIGS. 7A and 7B  denotes elapsed time from the time when nonlinear distortion compensation by the controller  50  based on the result of detection by the nonlinear effects monitor  40  is started. The axis of ordinates in  FIG. 7A  denotes the signal quality and that in  FIG. 7B  denotes the nonlinear effects monitor amount. 
         [0066]    It is found from  FIGS. 7A and 7B  that the signal quality is converged on an expected signal quality as the nonlinear effects monitor decreases over time. Accordingly, nonlinear distortion can be detected using the nonlinear effects monitor  40  according to the present embodiment. Nonlinear distortion can be compensated for based on the result of detection. Thus, the nonlinear distortion can be compensated for with high accuracy. 
         [0067]    To increase a calculation speed of the nonlinear effects monitor  40 , the main signal processor  30  may include a noise addition mechanism. When the error correction calculator  35  is separated into two stages, an error correction number may be extracted from a first-stage decoder or a second-stage decoder, or may be generated by combining both of error correction numbers. As for an example of the two-stage structure, the error correction calculator  35  may include a soft-decision forward error correction (FEC) device and a hard-decision FEC circuit. 
         [0068]    In the present embodiment, a Q value is used as a signal quality index. The signal quality index is not limited to the Q value. For example, a received signal spread in the vicinity of each signal constellation point may be used.  FIG. 8A  illustrates received signal spreads in the vicinities of signal constellation points. Referring to  FIG. 8A , the axis of abscissas indicates the I phase and the axis of ordinates indicates the Q phase. The signal constellation points have four phases (π/4, 3π/4, 5π/4, 7π/4). The distribution of a received signal in the vicinity of each signal constellation point is expressed by the following expression based on Gaussian distribution. 
         [0000]    
       
         
           
             
               
                 
                   
                     p 
                      
                     
                       ( 
                       x 
                       ) 
                     
                   
                   = 
                   
                     
                       1 
                       
                         σ 
                          
                         
                           
                             2 
                              
                             π 
                           
                         
                       
                     
                      
                     
                       exp 
                       ( 
                       
                         
                           - 
                           
                             x 
                             2 
                           
                         
                         
                           2 
                            
                           
                             σ 
                             2 
                           
                         
                       
                       ) 
                     
                   
                 
               
               
                 
                   ( 
                   7 
                   ) 
                 
               
             
           
         
       
     
         [0069]    As expressed in Expression (7), the parameter σ denotes an amount of fluctuation (signal quality) around each signal constellation point. As disclosed in Theodore S. Rappaport, “Wireless Communications-principles &amp; Practice”, Pearson Education, Inc, 1996, the probability that a fluctuation exceeds a value X 0  (Q value−signal quality) can be expressed by the following expressions. The fluctuation X 0  can be expressed as illustrated in  FIG. 8B . 
         [0000]                    Q        (   z   )       =       ∫   z   ∞            1       2      π              exp   (       -     y   2       2     )             y                 (   8   )                 y=x/σ   (9)
 
         [0000]        z=x   0 /σ  (10)
 
         [0070]    When an optimum signal identification point Z is selected, a Q value is combined with a signal error rate (BER) based on the relation of Q=√2erfc −1 (2×BER), as disclosed in Takaya Yamamoto, “Hikari Fiber Tsushin Gijutsu [Optical Fiber Communication Technology]”, Nikkan Kyogo Shimbun, Ltd., 1995. Accordingly, a fluctuation around each signal constellation point, a Q value, and a signal error rate can be used as different indices substantially indicating the same. 
       Second Embodiment 
       [0071]      FIG. 9A  illustrates an optical receiver  103   a  according to a second embodiment. The optical receiver  103   a  is a polarization diversity optical receiver. Referring to  FIG. 9A , the optical receiver  103   a  includes a coherent optical front-end  10   a  instead of the coherent optical front-end  10  as illustrated in  FIG. 2A . 
         [0072]      FIG. 9B  illustrates the coherent optical front-end  10   a . Referring to  FIG. 9B , an optical signal is input to a polarization beam splitter  15 . The polarization beam splitter  15  splits the optical signal into two optical signals in two polarization directions. A polarization beam splitter  16  splits an oscillation optical signal of the local optical oscillator  14  into two optical signals in two polarization directions. The 90° hybrids  12 , the O/Es  13 , and the ADCs  20  convert the polarized optical signals to I-phase and Q-phase digital electrical signals. The signals are input to the main signal processor  30 . 
         [0073]    In the optical receiver  103   a , the nonlinear effects monitor  40  according to the first embodiment is provided for each polarization, so that a nonlinear effects monitor amount for each polarization can be acquired. Thus, nonlinear distortion can be compensated for with high accuracy. 
       Third Embodiment 
       [0074]      FIG. 10A  illustrates an optical receiver  103   b  according to a third embodiment. The optical receiver  103   b  is a self-coherent optical receiver. Referring to  FIG. 10A , the optical receiver  103   b  includes a coherent optical front-end  10   b  instead of the coherent optical front-end  10  and includes a main signal processor  30   b  instead of the main signal processor  30  as illustrated in  FIG. 2A . 
         [0075]      FIG. 10B  illustrates the coherent optical front-end  10   b . Referring to  FIG. 10B , an optical signal input to the optical receiver  103   b  is split into two signals by a beam splitter  17 . One of the split optical signals is further split by a beam splitter  18 . Optical signals obtained by the beam splitter  18  are supplied to delay interferometers  19 , respectively. One of the delay interferometers  19  allows the optical signal to cause self delay interference, thus extracting an I-phase signal included in the optical signal. For example, an input signal is split into two signals and one of the signals is delayed by one bit and is allowed to interfere with the other signal, so that an I-phase signal can be extracted. The other delay interferometer  19  allows the optical signal to cause self delay interference, thus extracting a Q-phase signal included in the optical signal. The optical signals output from the delay interferometers  19  are input to the O/Es  13 . 
         [0076]    The other split optical signal obtained through the beam splitter  17  is directly input to the O/E  13  without bypassing the delay interferometer  19 . The functions of the O/Es  13  and the ADCs  20  may be the same as those in the first embodiment. Digital electrical signals output from the ADCs  20  are input to a main signal processor  30   b.    
         [0077]    The main signal processor  30   b  includes an electric-field reconstruction circuit  36  at the preceding stage of the equalizer  31  and includes a multi-symbol phase estimation (MPSE) circuit  37  instead of the carrier phase recovery circuit  33  as illustrated in  FIG. 2A . The electric-field reconstruction circuit  36  performs a process of reconstructing a received complex optical electric field from received signal information and outputs the resultant signal as I-phase and Q-phase signals to the equalizer  31 . Each of the equalizer  31  and the sub equalizer  32  compensates for distortion of the input electrical signal and outputs the resultant electrical signal to the MPSE circuit  37 . The MPSE circuit  37  performs multi-symbol phase estimation on the input signal and outputs the resultant signal to the identification calculator  34 . 
         [0078]    In the present embodiment, a nonlinear effects monitor amount can be acquired using signals upstream and downstream of the identification calculator  34  and an error correction number in accordance with a method similar to the first embodiment. Thus, nonlinear distortion can be compensated for with high accuracy. 
       Fourth Embodiment 
       [0079]      FIG. 11A  illustrates an optical receiver  103   c  according to a fourth embodiment. The optical receiver  103   c  is a polarization diversity self-coherent optical receiver. Referring to  FIG. 11A , the optical receiver  103   c  includes a coherent optical front-end  10   c  instead of the coherent optical front-end  10  and includes a main signal processor  30   c  instead of the main signal processor  30  as illustrated in  FIG. 2A . 
         [0080]      FIG. 11B  illustrates the coherent optical front-end  10   c . Referring to  FIG. 11B , an optical signal input to the optical receiver  103   c  is split into two optical signals in two polarization directions by the polarization beam splitter  15 . The split optical signals are processed through the delay interferometers  19 , the O/Es  13 , and the ADCs  20  according to the third embodiment and are then input to the main signal processor  30   c.    
         [0081]    The main signal processor  30   c  includes two electric-field reconstruction circuits  36  at the preceding stage of the equalizer  31  and includes the MPSE circuit  37  instead of the carrier phase recovery circuit  33  as illustrated in  FIG. 2A . Digital electrical signals corresponding to the two split optical signals in the two polarization directions are input to the electric-field reconstruction circuits  36 , respectively. 
         [0082]    Each electric-field reconstruction circuit  36  performs a process of reconstructing a complex optical electric field in a manner substantially similar to the third embodiment and outputs the resultant signal to the equalizer  31 . The other functions and components in the fourth embodiment may be the same as those in the third embodiment. 
         [0083]    In the present embodiment, a nonlinear effects monitor amount can be acquired using signals upstream and downstream of the identification calculator  34  and an error correction number in accordance with a method similar to the first embodiment. Thus, nonlinear distortion can be compensated for with high accuracy. 
         [0084]    While the embodiments have been described in detail, it should be understood that the present invention is not limited to any specific embodiment and various changes and modifications of the embodiments may be made within the scope and spirit of the present invention defined in the appended claims. 
         [0085]    The nonlinear distortion detecting module, the optical receiver, the optical transmission system, and the method for detecting nonlinear distortion disclosed in this specification can detect nonlinear distortion that that can be used for nonlinear distortion compensation.