Abstract:
A method and test setup for reducing the RF power requirement in active load pull uses impedance tuners between the output of the test transistor and the active RF power injection network. The active network uses either a closed loop (active load) configuration or an open loop network employing split or synchronized signal sources. The impedance tuners are wideband (fundamental) tuners, harmonic rejection tuners or multi-harmonic tuners. A 7:1 transforming ratio of the tuners represents the best compromise between power matching and tuner loss, yielding a reduction of 11dB in power requirements from the active load or the synchronized source(s); if only the fundamental signal is injected at the output of the DUT, a multi-harmonic tuner or a harmonic rejection tuner is used for independent harmonic tuning; if multiple harmonic signals are injected, a multi-harmonic tuner is used and creates passive harmonic loads while reducing at the same time the power requirements from the harmonic injection sources.

Description:
PRIORITY CLAIM 
     Not applicable 
     CROSS-REFERENCE TO RELATED ARTICLES 
     
         
         
           
             [1] Product Note 42, “Active Modules for harmonic load pull measurements”, Focus Microwaves, April 1997 
             [2] U.S. Pat. No. 7,135,941, Triple probe automatic slide screw load pull tuner and method. 
             [3]U.S. Pat. No. 6,297,649, Harmonic rejection load tuner. 
             [4] Product Note 79, “MPT, a universal Multi-Purpose Tuner”, Focus Microwaves, October 2004. 
             [5] Datasheet, XIANGYU COMMUNICATION TECHNOLOGY CO, LTD. 
             [6] “Overcoming Nonlinear Measurement Challenges”, Application Note, Tektronix Corp. 
             [7] Application Note 6, “Load Pull Measurements on very low impedance transistors”, Focus Microwaves, November 1993, Tables 4a, 4b, frequency points: 1, 2, 3 GHz. 
             [8] “A new load-pull characterization method for microwave power transistors”, Y. Takayama, 1976 IEEE-MTT-S symposium, pp. 218-220, June 1976. 
             [9] Cascading Tuners For High-VSWR And Harmonic Load Pull, Note 5C-081, Maury Microwave Corp. 
             [10] “Recent technological advances for high-VSWR and harmonic load pull”, Ferrero et al. 
             [11] RFS, Diplexer-Triplexer Datasheet. 
             [12] Wikipedia, “Quarter wave impedance transformer”. 
             [13] “Understanding power splitters”, AN-10-006, Mini Circuit Laboratories 
             [14] “How does a tuner work?”, Website, Focus Microwaves Inc, http://www.focus-microwaves.com/template.php?unique=157 
             [15] Harmonic Impedance Tuner with four wideband probes and method; U.S. patent application Ser. No. 12,457,187; Jun. 3, 2009 
           
         
       
    
     STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPEMENT 
     Not applicable 
     REFERENCE TO SEQUENCE LISTING, A TABLE, OR A COMPUTER PROGRAM LISTING COMPACT DISC APPENDIX 
     Not applicable 
     BACKGROUND OF THE INVENTION 
     This invention relates to microwave load pull testing of transistors and other active devices using impedance tuners. 
     Modern design of high power microwave amplifiers and oscillators, used in various communication systems, requires accurate knowledge of the active device&#39;s (microwave transistor, DUT) characteristics. In such circuits, it is insufficient for the transistors, which operate in their highly non-linear regime, close to power saturation, to be described using non-linear numeric models. 
     A popular method for testing and characterizing such microwave components (transistors) in the non-linear region of operation is “load pull”. Load pull is a measurement technique which allows changing the impedance seen by the DUT in a controlled manner, in order to explore the DUT&#39;s performance under such impedance conditions and find the optimum environment conditions for certain performance goals. One widespread technique employs passive microwave impedance tuners and other microwave test equipment for this task; said microwave tuners are used in order to reflect part of the outgoing power from the DUT and thus manipulate the microwave impedance conditions under which the Device Under Test (DUT, or transistor) is tested ( FIG. 1 ). 
     DESCRIPTION OF PRIOR ART 
     Passive impedance tuners can reach reflection factors |Γ turner | of the order of 0.95, corresponding to impedances of 2.4Ω. The insertion loss between DUT and tuner reduces the available tuning range at the DUT reference plane. This is shown in  FIGS. 2 and 3 . Consequently, load pull systems using passive impedance tuners have tuning range limitations. In order to compensate for said limitations “active” techniques have been proposed, in which RF power is, artificially, injected into the DUT in order to replace the power simply reflected at a passive load. 
     Such load pull techniques use the so called “virtual load” method. The virtual load method consists in injecting into the output of the DUT RF power at the same frequency as traversing the DUT from the input to the output, but with controlled phase and amplitude. Knowing that the reflection factor “seen” by the DUT is equal to the ratio of returned (normally “reflected”) power wave (Pr) to primary output power wave (Pout): (|Γ|=sqrt (Pr/Pout)), it is understood that by controlling the phase and amplitude of Pr we can control T. There are a number of ways to do this [10]: One is to use an “active load” technique ( FIG. 4 ). In this case part of the outgoing power Pout is split into a directional coupler, filtered, phase adjusted and amplified and re-injected into the output port of the DUT, which will “see” a complex load Γ as described above. Another method is to use a split signal [8] or two synchronized RF sources or one split source [6], one to inject RF power into the input of the DUT and one into its output ( FIG. 5 ). Since the sources are synchronized they can generate a coherent signal and if we adjust the relative phases between sources and the amplitude of the second source, then the DUT will, again, see at its output port a “virtual” load Γ, as described earlier. 
     The alternative open loop active load pull approach to nonlinear measurements in  FIG. 5  uses different signal sources to inject power into either the source or load side of the DUT. The open loop system absorbs the signal that is generated by the device under test (DUT) and injects back into the device a signal that is generated by an independent coherent but amplitude and phase modulated source. The power amplifier bandwidth should be sufficiently large to cover all harmonic frequencies at which the impedance control is required. Because of high linearity requirements it is often necessary (at higher power levels) to implement the amplification over the required number of harmonics through use of several narrowband power amplifiers operating at center frequencies around the fundamental and harmonic frequencies. 
     As the state of the art stands today, it is claimed that in both cases of “active signal injection” there is no need for an impedance tuner in order to manipulate the impedance (reflection factor) presented to the DUT. Since the re-injected power Pr can be amplified to levels equal (or even exceeding) Pout, the reflection factor presented to the DUT can be equal to (or even higher than) 1. Of course it does not make sense presenting |Γ|&gt;1 to the DUT, since |Γ|&gt;1 corresponds to a negative real part R of the complex impedance Z=R+jX, and it would be impossible to deliver RF power into a negative resistance. Getting very close to |Γ|=1 is, however, an attractive goal, since the internal impedance of power transistors R 1  is very close to (or even lower than) 1Ω. For impedances with zero imaginary parts (an often encountered situation) the simple relation |Γ|=(50Ω−R 1 )/(50Ω+R 1 ) can be used. If the output impedance of the DUT is R 1 =1Ω, then the corresponding reflection factor Γ=0.96 ( FIG. 6 ). 
    
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
       The invention and its mode of operation will be more clearly understood from the following detailed description when read with the appended drawings in which: 
         FIG. 1   a  depicts prior art: a wideband load pull test system for testing transistors using passive wideband tuners.  FIG. 1   b  depicts prior art: the position in the setup where insertion loss affects the reflection factor of the tuner, seen by the DUT. 
         FIG. 2  depicts prior art: the effect of insertion loss of the section between tuner and DUT on the effective tuning range of the tuner. 
         FIG. 3  depicts prior art: an “active loop” load pull test system, where a sample of the outgoing signal is amplified and phase adjusted to be re-injected into the output of the DUT. 
         FIG. 4  depicts prior art: an active injection load pull test system using direct injection from two synchronized sources at the same fundamental frequency using a 50 ohm circulator. 
         FIG. 5  depicts prior art: a commercial load pull test system [6] using an injection arbitrary waveform source with the signal split between the input and output of the DUT, again using a 50 ohm circulator. 
         FIGS. 6   a, b  depict prior art, showing the concept of injecting power from a generator into a mismatched load and the effect of a transformer. 
         FIG. 7  depicts prior art: an active fundamental frequency (fo) load pull test setup, where a second signal source generates a signal at fo which is injected into the output of the DUT and uses a 50 ohm circulator and a passive (wideband) impedance tuner for reducing the power requirement of the injection power amplifier. 
         FIG. 8  depicts the same, prior art, setup as  FIG. 7 , except the circulator is replaced by a directional coupler (DC) in order to inject the signal from source  2  into the DUT. 
         FIG. 9  depicts a new hybrid active injection load pull system using a second coherent source and a passive harmonic tuner, wherein tuning with the harmonic tuner at the fundamental frequency (fo) pre-matches the fundamental impedance and reduces injected power, and, simultaneously, tuning at the harmonic frequencies ( 2   fo ) and ( 3   fo ) is created using passive harmonic tuning only. 
         FIG. 10  depicts a setup as in  FIG. 9 , except active injection tuning, using passive pre-matching, is at the fundamental frequency (fo) and passive tuning at three harmonic frequencies ( 2   fo ,  3   fo  and  4   fo ) using a four probe harmonic tuner [15]. 
         FIG. 11  depicts prior art: output section of pre-matching test fixture using a Lambda/4 micro-strip transformer with a characteristic impedance Zc&lt;Zo. 
         FIG. 12  depicts a hybrid active injection harmonic load pull test setup using three coherent signal sources injecting power at the DUT output at the fundamental (fo) and two harmonic frequencies ( 2   fo ,  3   fo ), and a harmonic tuner used to pre-match all frequencies (fo,  2   fo  and  3   fo ) in order to reduce injected power at all frequencies. 
         FIG. 13  shows the mechanism of passive pre-matching at  3  harmonic frequencies simultaneously (hybrid harmonic tuning), whereby Γ tun (nfo) is the passive reflection factor generated by the harmonic tuner and Γ inj (nfo) is the reflection factor created by active power injection. 
         FIG. 14  depicts a hybrid system like in  FIG. 12 , wherein the output power amplifier is not included. 
         FIG. 15  depicts a hybrid system like  FIG. 12 , wherein only two injection sources are used and the harmonic tuner is replaced by a cascade of two wideband tuners, which has the capacity to perform harmonic tuning at two frequencies. 
         FIG. 16  depicts a hybrid setup as in  FIG. 12  wherein the harmonic tuner is replaced by a cascade of three wideband tuners which has the capacity to perform harmonic tuning at three frequencies. 
         FIG. 17  depicts a hybrid setup, wherein a single output injection source and a wideband tuner are used to pre-match at the fundamental frequency (fo) and wherein a passive harmonic rejection tuner [3] is used to generate high reflection at the second and third harmonic frequencies. 
         FIG. 18  depicts a hybrid harmonic load pull test setup, wherein the DUT is pre-adapted by mounting in a pre-matching test fixture, which uses a lambda/4 transformer, and a two frequency harmonic tuner is used to pre-match the fundamental and first harmonic frequencies (fo,  2   fo ), injected into the DUT output from two coherent injection sources. 
         FIG. 19  depicts a hybrid setup wherein a cascade of two wideband tuners is used to pre-match an already pre-adapted DUT in a test fixture which uses a lambda/4 transformer. 
         FIG. 20  depicts the typical frequency response of the various types of probes used in wideband slide screw tuners. 
         FIG. 21  depicts a harmonic hybrid setup, wherein a (typically) 3 dB attenuator is used to reduce the residual reflection of a wideband circulator outside its operation band, without increasing dramatically the requirement for injected power. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The present invention is about reducing the power requirements in the circuit of the active power injection into the DUT. As already shown an active injection system, consisting of an “active loop” or “two sources”, uses a (typically) 50Ω generator (the circulator in  FIG. 3  or the directional coupler in  FIG. 8  are both components with 50Ω internal impedance) to inject RF power into the output port of the DUT, which is a low value impedance of typically 0.5 to 2Ω.  FIG. 6  illustrates the base of the power transfer calculations. In case of  FIG. 6   a  (no transformer) the power delivered from the source Vo to the load R 1  is: P 1 =Vo 2 *R 1 /(R 1 −Ro) 2 . In the case of  FIG. 6   b  (transformer 1:n included) the power delivered from the source Vo to R 1  is: P 1 =Vo 2 *n 2 *R 1 /(Ro−n 2 *R 1 ) 2 . The ratio of power delivered to R 1  directly or through a transformer is then: P 1 (direct)/P 1  (transformed)=(Ro÷n 2 *R 1 ) 2 /(n 2 *(R 1 −Ro) 2 ). In a typical case of R 1 =1Ω, Ro=50Ω, n=7, this ratio becomes: P 1 .direct/P 1 .transf=1/13=0.077. In short, adding a 7:1 transformer reduces the power requirement by a factor of 13 (=11.14 dB). If R 1 =0.55, and with a transforming ratio of 10:1, the power ratio becomes: P 1 .direct/P 1 .transf=1/25=0.04 (the power needed becomes 25 times less, or 14 dB). These are very important savings in power requirement from the RF power amplifiers, since said amplifiers need to operate under very linear conditions in order not to create spurious signals in the measurement. 
     Transformers, including impedance tuners, are not lossless. Depending on the transforming ratio the loss own to the transformer may be significant up to a level in fact canceling the power transfer gain obtained due to their use. Typically microwave transformers have loss of the order of 0.1 to 3 dB, depending on their manufacturing procedure, frequency and transforming ratio, up to transforming ratios of 10:1. Microstrip, fixed, transformers ( FIG. 11 ) have low loss, of the order of 0.5-2 dB, again for typical transforming ratios up to 10:1. Impedance tuners, as proposed here, have loss between 0.5 and 2 dB depending on frequency and transforming ratio. At 2 GHz a 7:1 impedance transformer has less than 0.5 dB loss. This does not reduce significantly the advantage in power transfer gained by using said transformers in this application. 
     High power transistors are operated in their nonlinear regime, close to saturation. This is required in order to increase amplifier efficiency and battery life in mobile equipment, like mobile phones. Under these conditions the transistors generate harmonic signals. If said harmonic signals are not properly terminated at the output of the DUT, the efficiency and linearity of the amplifiers degrades. Thus the necessity of controlling the harmonic impedances at the output of the DUT as well as the main (fundamental) frequency (fo). Harmonic frequencies are named here as  2   fo ,  3   fo  etc. The corresponding impedances are Z( 2   fo ), Z( 3   fo ), etc. (or reflection factors Γ( 2   fo ), Γ( 3   fo ) etc.); noticing that Γ=(Z−Zo)/(Z+Zo), with Z=R+jX being the complex impedance and Zo=50Ω the typical characteristic impedance of microwave networks. Harmonic impedance control is possible using appropriate harmonic tuners [PHT, MPT patents and literature]. 
       FIGS. 9 ,  10 ,  12  and  14  to  19  show various possible configurations of active load pull test systems with harmonic impedance control. At each frequency present at the output of the DUT (fo,  2   fo ,  3   fo  etc.) it is possible to create a “real”, a “virtual” or a “combination” load impedance either using active injection from an external source or a real load using an impedance tuner or both ( FIG. 13 ). In  FIG. 13  the individual reflection factors at each harmonic frequency are shown to be the vector sum of a primary reflection factor Γ tun  coming from the impedance tuner and a secondary reflection factor Γ inj  coming from the external signal source through active injection network. The choice of method depends on the frequency (fo) itself and the characteristics of certain components, such as the circulator C ( FIG. 10 ) [5]. Circulators covering a full range of three harmonic frequencies are rare. They exist covering bandwidths between 2 and 6 GHz or 6 and 18 GHz, but no more. So, active injection must in certain cases be provided by a wideband coupler, such as a 3 dB coupler, DC,  FIG. 8 ). In this case the loss in power at the output is 3 dB (=half of the injected power reaches the DUT), but the gain in power transfer efficiency due to the transformer, in general, over-compensates for that. 
     Using multi-harmonic passive tuners [2, 15] in a load pull setup offers important advantages when seeking full harmonic impedance control: a) the harmonic impedances can be controlled, directly at the output of the DUT using said (passive) tuner. Only RF power at the fundamental frequency (fo) needs to be actively injected by the second source,  FIG. 9  (or the active loop,  FIG. 3 ), in which case the limited bandwidth of a circulator is not really a handicap. On the contrary, circulators having insertion loss of 0.5 dB or less [5] are clearly superior to wideband 3 dB couplers (which have 3 dB loss). The configuration of  FIG. 9  is then a superior solution to the configuration of  FIG. 12 ; the fact that insertion loss between DUT and tuner ( FIGS. 1   b ,  2 ) reduces the tuning range at the harmonic frequencies as well, is of lesser importance, since harmonic tuning has anyway a non-negligible but nevertheless secondary effect on DUT performance. 
     In case of  FIG. 12 , a wideband coupler is used, through which harmonic signals are injected in order to enhance harmonic tuning range at the DUT port, despite the coupler&#39;s at least 3 dB insertion loss, the multi-harmonic tuner can still be useful in pre-matching the high output impedance of the DUT to the 50Ω internal impedance of the coupler, as can be understood by comparing with  FIG. 13 . 
     So, either using active injection at the harmonics or not, a multi-harmonic tuner, allowing independent pre-matching between the DUT and the active load brings advantages in reducing the RF power required from said sources associated with the active load. Said RF power is provided by complex and costly power amplifiers, which can be downsized. In certain (higher) frequency ranges, such as fo=10 GHz (corresponding to harmonic frequencies  2   fo =20 GHz and  3   fo =30 GHz) said power amplifiers are rare, unstable and very expensive. The associated networks become cumbersome and sensitive. Such solutions are so impractical that, even if theoretically possible, they are rarely employed. The test setups proposed here have key advantages, in that they are stable, cost effective and easy to calibrate and operate. 
     Harmonic tuning using wideband multi-harmonic tuners consists in searching for simultaneous impedance solutions at the harmonic frequencies among a large amount of possible alternatives [4]. The load impedance Γ c  at the output of said tuners affects the capacity of tuning at the DUT output (Γ L ) all over the Smith chart ( FIG. 21 ). Circulators have a very low reflection factor inside their operation bandwidth, but have important reflection outside of it. In this case the load reflection factor Γ c  presented to said harmonic tuner must be reduced into a manageable level over the whole frequency bandwidth; this can be done using a moderate size passive attenuator;  FIG. 21  shows the use of a 3 dB attenuator. 3 dB is a good compromise, because it reduces the reflection factor by 6 dB (3 dB in each direction) but the reverse injected power at the fundamental frequency (fo) by only 3 dB. If for instance the circulator has a high reflection factor of 0.6 (Return Loss=−4.44 dB) outside its operation band and is damped by a 3 dB attenuator, then said harmonic tuner will “see” a residual reflection reduced by 6 dB: Γc=0.3 (=−10.5 dB), which will allow proper harmonic tuning over most of the Smith chart at the DUT output reference plane. If 3 dB is insufficient for adequate harmonic tuning, then higher attenuation can be used, knowing that this will increase the need for higher power from the source and the associated amplifiers. 
     A simpler (and cheaper) alternative would be, of course, to use a micro-strip transformer ( FIG. 11 ). Such a transformer can be designed to adapt, with little loss, high impedance to low impedance. In the example of  FIG. 11 , said micro-strip transformer has a quarter lambda long section with a characteristic impedance Zc of 7Ω. A basic formula [12] states that the impedance seen on both sides of such a transformer follows the rule: Zc 2 =Zo*Z 1 . With Zc=7Ω and Zo=50Ω, then the impedance seen into the transformer by the DUT is Z 1 ≈1Ω. This means the DUT will be quasi power matched. The disadvantages of fixed micro-strip transformers, beyond the fact that they cannot be used for on-wafer testing, is a limited bandwidth and the fact that they will also transform the odd harmonic impedances (Z( 3   fo ), Z( 5   fo ), etc.) into the same basic area of the Smith chart as the fundamental impedance Z(fo); this is because λ/4, 3λ/4, 5λ/4 etc. sections of transmission line turn the angle of the reflection factor by 180 degrees, thus transforming a high impedance into a low impedance, whereas sections of λ/2, λ, 3/2λ, etc. turn the angle of the reflection factor by 360 degrees, thus maintaining the impedance level as before the transformation [7]; in reference [7] tables  4   a  and  4   b  show measured s-parameters in amplitude and angle format (|S11|&lt;S11|S12|&lt;S12|S21|&lt;S21|S22|&lt;S22) of the input and output block of a test fixture, each having a quarter lambda micro-strip transformer, centered at fo=1 GHz, on it; the s-parameters of said sections at fo=1.00 GHz,  2   fo =2.00 GHz and  3   fo =3.00 GHz show that at fo (1.00 GHz) and  3   fo  (3.00 GHz) the reflection (|S11|, |S22|) of each said block is high (˜0.65 to 0.7) and at  2   fo  (2.00 GHz) it is low (0.047); this proves the above claim that a lambda quarter transformer is “transparent” at  2   fo  and by extrapolation  4   fo ,  6   fo  etc. and creates high reflection at odd harmonic multiples ( 3   fo ,  5   fo  etc.). The optimal harmonic impedances at the odd harmonic frequencies is not the same as at the fundamental frequency and therefore tuning capability in a different area of the Smith chart is irrevocably reduced. In summary lambda quarter transformers are transparent (albeit with some insertion loss) for even harmonic frequencies in which case the impedances at those frequencies (Z( 2   fo ), Z( 4   fo ), etc. see reference [6]) can be tuned with small disturbance through the lambda quarter transformer, by power injection or multi-harmonic tuner, but with no other advantage ( FIG. 18 ). 
       FIGS. 14 and 16  show the configuration where three synchronized sources are used to inject harmonic power into the transistor output. Harmonic power generated by the transistor is, in general, lower than the fundamental power, except in cases of extreme nonlinearities. Normally harmonic power is 10 to 20 dB lower than the fundamental signal. Therefore the injected power to be provided from the synchronized sources is also lower. The fundamental and harmonic signals are injected through a wideband directional coupler (DC). The multi-harmonic tuner facilitates the power transfer from the coupler port to the DUT port. In terms of reflection factor this can be seen as “assisted” reflection factor ( FIG. 13 ). It is an efficient way to increase the actual reflection factor at the DUT output port. 
     The signals generated by the two or three synchronized sources at the output of the test setup must be combined and injected into the output port of the DUT. An efficient way to do this is using a frequency multi-plexer, a Triplexer [11] in case of three frequencies ( FIG. 12 ) or a Diplexer in case of two frequencies [11]. In case of only two frequencies (fo and  2   fo ) the requirements on the circulator as well as the pre-matching harmonic tuner are simpler. Circulators with an octave bandwidth are easier to have and impedance tuners with two instead of three independent probes can be used. 
     Frequency multi-plexers (Di- and Tri-plexers, [11]) are frequency selective and have limitations regarding RF power. An alternative is to use simple wideband power combiners ( FIGS. 14 ,  15 ,  16 ), [13], with an associated wideband circulator or directional coupler. In each case the insertion loss is different, but limitations in circulator bandwidth [5] may impose the use of wideband directional couplers. 
     Harmonic tuning using impedance tuners is possible, as mentioned before, using either multi-probe impedance tuners which comprise two or more wideband probes in the same low loss transmission line [2, 4]. Using a cascade of two or more independent wideband tuners can achieve the same result, albeit with lower reflection factor and accuracy performance, due to insertion loss in each cascaded tuner and mismatch loss of each tuner&#39;s adapters [9]. Such configurations are shown in  FIGS. 15 ,  16  and  19 . 
     In the setup of  FIG. 15  the two cascaded wideband tuners can be selected such as to have a low pass behavior allowing the fundamental frequency (fo) to pass through unaffected and create reflection only for frequencies in the range of the harmonics ( 2   fo  and  3   fo ). This can be done by proper selection of the tuner probes [14],  FIG. 20 . In this case said cascaded tuners can create harmonic impedances only at  2   fo  and  3   fo  and allow active power injection pass through to the DUT. This configuration is recommended when the bandwidth of the circulator is insufficient to handle the harmonics, as is the case for fundamental frequencies below 2 GHz [5]. 
     Another method consists in using harmonic rejection tuners [3],  FIG. 17 ; harmonic rejection tuners use narrow-band resonator heads and generate high reflection at the harmonic frequencies, without the capability of adjusting the amplitude of the reflection factor. Therefore said harmonic rejection tuners are not suitable for pre-matching the harmonic signals coming from the active load, which in this case injects only fundamental RF power and not harmonic signals. Said harmonic rejection tuners create only high reflection factors with phase control at the harmonic frequencies but allow the fundamental signal power (which is the more critical one) to traverse with very little insertion loss; said fundamental power is traversing from the active load and is pre-matched by an additional wideband tuner, which is inserted between the output of the harmonic rejection tuner and the circulator. 
     It has been found that the ratio of tuner loss versus improvement of power transfer degrades significantly for transforming ratios larger than 10:1. This means that, when an impedance tuner is used to pre-match the active load source(s) the recommended transforming ratio is ˜7:1. At this level the tuner loss is low and improvement of RF power transfer is high. The typical overall gain in power transfer exceeds 10 dB (a factor of 10 in RF power), which is very significant: the cost of a 100 Watt amplifier (when available) can be 5 to 10 times higher than that of a 10 Watt amplifier. In order to match the RF power delivered by a power transistor (meaning to generate a reflection factor close to 1) we need to inject into its output port almost as much power as the transistor itself generates. A (typical) 10 Watt transistor would need, if pre-matching is used roughly a 15 Watt amplifier, assuming typical insertion and coupling losses in the injection path; the same transistor would require a 150 Watt amplifier if RF power is injected directly, without pre-matching. The same reasoning exists at the harmonic frequencies, albeit the powers involved are typically 10 to 20 dB (=10 to 100 times) lower, but the frequencies are also higher and so is the cost of the corresponding amplifiers. 
     The concept of using impedance tuners, partly combined with fixed matching networks, in order to improve the RF power transfer and reduce power requirements of an active injection network into the output port of a power transistor in a load pull system can be materialized in a number of possible ways; several such configurations with practical significance have been outlined in this invention; this shall not limit the claims to obvious combinations and variations of this basic concept.