Abstract:
The present invention relates to a vehicle mountable satellite antenna as defined in the claims which is operable while the vehicle is in motion. The satellite antenna of the present invention can be installed on top of (or embedded into) the roof of a vehicle. The antenna is capable of providing high gain and a narrow antenna beam for aiming at a satellite direction and enabling broadband communication to vehicle. The present invention provides a vehicle mounted satellite antenna which has low axial ratio, high efficiency and has low grating lobes gain. The vehicle mounted satellite antenna of the present invention provides two simultaneous polarization states. In one embodiment, an inverted L-shaped waveguide has a first wall extending vertically downward from a top surface. The top surface can include a ridge portion. The top surface includes a plurality of radiating elements for forming a radiating surface.

Description:
BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The invention relates to vehicle mounted satellite antennae. More particularly, the invention relates to a slotted waveguide planar vehicle mounted satellite antenna with simultaneous dual polarization states, which can employ a hybrid electronic and mechanic steering mechanism, and is operable while the vehicle is in motion. 
   2. Related Art 
   It has long been known how to mount a satellite antenna (dish) atop a vehicle for purposes of communicating with a geostationary or other type of satellite. The initial applications for mounting a satellite dish on a vehicle were military communication and remote television news broadcasting. Consequently, the first methods of mounting a satellite dish included a telescoping mast which was hingedly coupled to the vehicle. When the vehicle was in motion, the mast would be retracted and folded with the satellite dish lying end up on the roof or a side wall of the vehicle. The dish would be deployed only when the vehicle was stationary. Such a deployable vehicle mounted satellite dish is disclosed in U.S. Pat. No. 5,961,092 to Coffield. Until recently, no vehicle mounted satellite antennae were operable while the vehicle was in motion. The relatively large size of a conventional satellite dish antenna presents significant wind resistance if deployed on a vehicle in motion. This wind resistance adversely affects the operation of the vehicle and subjects the satellite dish to potential wind damage. Moreover, satellite dishes must be accurately aimed at a satellite within a relatively narrow aperture or “look window”. In order to operate a satellite dish mounted on a vehicle in motion, it would be necessary to constantly re-aim the dish in order to maintain communication with the satellite. 
   Recently, satellite antennae have been developed which may be deployed on a vehicle and operated while the vehicle is in motion. Such antennae are disclosed in U.S. Pat. No. 5,398,035 to Densmore et al., U.S. Pat. No. 5,982,333 to Stillinger, and U.S. Pat. No. 6,049,306 to Amarillas. These antenna systems generally include a satellite antenna of reduced size and a solenoid system for aiming the antenna. The solenoid system is coupled to a feedback system and/or vehicle motion detectors in order to automatically re-aim the antenna as the vehicle is in motion. In order to reduce aerodynamic drag and protect the antenna from wind damage, an aerodynamic radome is often used to cover the antenna. 
   Vehicle mounted satellite antennae which are operable while the vehicle is in motion, can provide one-way or two-way satellite communications. Some applications for such antennae include satellite television reception, telephony in remote locations where cellular telephone service is unavailable, and broadband data communications. The application of television reception may be advantageously applied in common carrier transportation such as long distance buses, in recreational vehicles including boats, and in the rear seats of family mini-vans. The application of remote telephony may be applied in the same situations as well as in various other governmental and commercial settings. The application of broadband data communication may also be applied in many personal, commercial, and governmental settings. 
   Broadband satellite communication, such as television reception or broadband data communication requires a high gain antenna with high cross-polarization isolation and low signal sidelobes. Satellite antenna gain is proportional to the aperture area of the reflector. Stationary satellite antennae typically utilize a circular parabolic reflector. Reflector type of satellite antennae designed for use on a moving vehicle is difficult to achieve low profile. In order to maintain gain, these low profile antenna are short but wide so that the overall aperture area is kept high. However, this design strategy only works to a point. When the width to height ratio exceeds a certain value such as 2, the efficiency of the antenna is adversely affected. The presently available vehicle mountable dish reflector type of satellite antennas, for commercial and personal use, are no shorter than approximately fifteen inches in height. A mobile satellite antenna produced by Audivox Corp. (MVSTS Satellite TV System) provides four circular Casegrain dish reflector antennas positioned along a horizontal axis perpendicular to the direction of antenna aiming. The signals received by the four dish reflectors are combined in phase to achieve aggregate antenna gain. Since the signal arriving at the phase centers of the four reflectors with the same propagation delay, no phase shifters are required for this mobile satellite antenna. The use of four reflector dishes allow the width to height ratio to be stretched further, while maintaining the antenna efficiency. The overall height of this antenna including radome is approximately 9.5 inches, considerably reduced from the single reflector type of dish antenna. Another mobile satellite antenna produced by Titan corporation (DBS-2400 Low Profile Ku-Band Antenna System) uses four hemisphere Luneberg lens antennas positioned on top of a ground plate along a horizontal axis perpendicular to the direction of the antenna aiming. The signals received by four Luneberg lens antennas are combined. The use of the ground plate to create an image of the hemisphere antenna reduces the height of the Luneberg lens by half, to approximately 5 inches (including radom). Another approach described in U.S. Pat. Nos. 6,657,589 and 6,653,981 to Wang et al., is a linear cylindrical Casegrain reflector antenna with line source. Such antenna profile is also limited to approximately 5 inches without elongating the antenna length prohibitively. A common drawback of the antennas described above is that two dimensional mechanic movement and control is required to aim the antenna toward satellite. This makes the mechanic design complicated and reduces the reliability of the antenna system. Another drawback of these types of antennas is that the height of the antenna is still too large for esthetically mounting on top of the roof of the commercial vehicles such as mini-van or SUV (Suburban Utility Vehicle). Further, the Lunberg lens antenna approach is heavy and expensive. 
   Another approach for implementing the mobile satellite antenna is to employ a phased array antenna having a large number of antenna elements. An antenna aiming in the azimuth and elevation directions is achieved by passing the received signal from each antenna element through a phase shifter. The phase shifter rotates the phases of the signals received from all antenna elements to a common phase before they are combined. While such antennas can be implemented with a very low profile, the large number of microwave processing elements such as amplifiers and phase shifters used in the electronic beam forming network results in high implementation cost, preventing mass volume commercial use. One of such antenna was published by V. Peshlov et al. of Sky Gate BG, IEEE 2003, Phased-array antenna conference. 
   U.S. patent application Nos. 2003/0083063, 2003/0080907 and 2003/008098 describe an antenna mounted on a horizontal platform, which is rotatable to adjust the antenna beam in the azimuth direction driven by a motor, and is also capable of steering the antenna beam in the elevation direction through an electronic beam forming network. 
   Waveguide antennas are typically less than one wavelength in height and provide signal combining along the waveguide longitudinal axis. Many forms of waveguides can be used for microwave energy transmission. Rectangular waveguides have currents flowing on its interior wall and interrupting those currents by cutting through the waveguide wall can cause radiation into the exterior. It is well known, and used, that a radiating aperture is achieved when that aperture is approximately one-half free space wavelength long and one twentieth of a wavelength wide is cut through the broad wall of that waveguide. The aperture is widely described as a “slot” through the waveguide wall. Locating such a slot at various positions on the waveguide wall achieves varying degrees of excitation of microwave fields emanating from the slot. The microwave fields from the simple slot are characterized as being linearly polarized microwave fields. 
   Many applications for field radiating structures require that the radiated fields have the property of being circularly polarized. A widely used technique for producing a circular polarized radiating element is the cutting of a pair of slots through the broad wall of a rectangular waveguide. The two slots are typically caused to cross each other at ninety degrees to each other, and at the center of each slots length. Further, the crossed slot is normally placed on a line that is parallel to the waveguide axis and is a distance of approximately one quarter of the waveguide width away from the waveguide axis. 
   U.S. Pat. No. 3,503,073 to James Ajioka et al., and subsequently in IEEE Transaction On Antenna and Propagation, March 1974, describes using a dual polarized slot radiators in bifurcated waveguide arrays. The radiating element is a pair of crossed slots in the narrow wall of a bifurcated rectangular waveguide that couples even and odd modes. One linear polarization is excited by the even mode, and the orthogonal linear polarization is excited by the odd mode. Alternatively, one circular polarization can be excited through one of the pair of waveguides, whereas, the other circular polarization can be excited through another waveguide in the pair. The above-described antenna design approach has the drawback of unequal propagation velocities of the even and odd mode within the waveguide which causes the even and odd beam to point at different direction. In order to equalize the two group velocities, very narrow compensating slits within the waveguide wall are used, which reduces the waveguide bandwidth and significantly complicates the manufacturing complexity. 
   Another antenna described in IEEE Transaction of Vehicular Technology, January 1999 by K. Sakakibara et al., employs X-shaped slot located in the broadwall of a rectangular waveguide, approximately halfway between the center line and the narrow wall, to form a two-beam slotted leaky waveguide array. The broad side width of rectangular waveguide is approximately half the waveguide, and the cross slot center is offset from the center of the waveguide toward the sidewall by approximately 90 mil. The slot spacing along the waveguide is 0.874 inch. Such waveguide spacing can result in grating lobe when the beam is steered to different elevation angle. At higher elevation angle, the grating lobe becomes comparable in strength to the main lobe, thereby reduces the antenna gain. A right-hand circular polarization can be achieved by feeding the waveguide from one end, whereas a left hand circular polarization can be achieved by feeding the waveguide from the opposite end. One disadvantage of this antenna is that the beam direction of the right-hand polarization antenna is different than the beam direction of the left-hand polarization antenna. As the user switches from one polarization to the other polarization, the antenna rotates in azimuth direction in order to refocus the antenna toward the satellite, resulting in temporary disruption of signal reception. The antenna described above is designed for a fixed elevation beam angle. 
   U.S. Pat. No. 6,028,562 to Michael et al. describes a planar array of waveguide slot radiators of parallel waveguides which couples the electromagnetic signal from alternating +45 degree and −45 degree radiating slots interfaced on top of the waveguide to the slots on the broadwall of the waveguides via cavities which serve as impedance matching network. In a corresponding U.S. Pat. No. 6,127,985 to Michael et al., a similar slotted waveguide structure is employed. A T-shaped ridge waveguide is employed to realize closely spaced waveguide slot radiator to provide simultaneous dual polarization and suppression of grating lobes. The Michael patents have the disadvantage of complicated manufacturing processing. In addition, the patents use a rear-fed waveguide combining structure, which is not intended for electronic beam steering. 
   Conventional systems have focused the antenna beam toward the satellite while vehicle is moving using a mechanic dithering approach. In this approach, the antenna is rotated in both azimuth and elevation by a small angle, such as a fraction of the antenna beamwidth, to slightly off-point the antenna beam in the left, right, up, and down directions. The mechanic dithering involves controlling a motor to move the antenna platform. This approach has the shortcoming of a slow response and inaccuracies in the mechanic movement require the use of motion sensors (such as gyro, accelerometer, or compass) to aiding the tracking thereby resulting in significant signal degradation. Electronic dithering is faster, but still subject to the similar problems of slow response. The motion sensors are expensive. 
   It is desirable to provide a vehicle mounted satellite antenna which has simpler mechanical control and more reliable design. 
   SUMMARY OF THE INVENTION 
   The present invention relates to a vehicle mountable satellite antenna as defined in the claims which is operable while the vehicle is in motion. The satellite antenna of the present invention can be installed on top of (or embedded into) the roof of a vehicle. The antenna is capable of providing high gain and a narrow antenna beam for aiming at a satellite direction and enabling broadband communication to vehicle. The present invention provides a vehicle mounted satellite antenna which has low axial ratio, high efficiency and has low grating lobes gain. The vehicle mounted satellite antenna of the present invention provides two simultaneous polarization states. 
   In one embodiment, the present invention provides a ridged waveguide instead of a conventional rectangular waveguide to alleviate the effects of grating lobes. The ridge waveguide provides a ridged section longitudinally between walls forming the waveguide. A plurality of radiating elements are formed in a radiating surface of the ridged waveguide. The use of a ridged waveguide reduces the width of the waveguide, and thus, the spacing between the antenna slots. This suppresses the strength of the grating lobe. In conventional approaches, the length between cross slots along the waveguide is approximately one waveguide. The resultant beam points upward in the plane orthogonal to the waveguide axis. The present invention reduces the length between cross slots along the waveguide to further suppress the grating lobe. This results in further beam tilting away from the plane orthogonal to the waveguide axis. However, as long as the beam can be pointed to highest required elevation angle, the beam tilting does not have adverse effects on the overall system performance. 
   In an alternate embodiment, an inverted L-shaped waveguide has a first wall extending vertically downward from a top surface. The top surface can include a ridge portion. The top surface includes a plurality of radiating elements for forming a radiating surface. 
   In one embodiment, a hybrid mechanic and electronic steering approach provides a more reasonable cost and performance trade-off. The antenna aiming in the elevation direction is achieved via control of an electronic beamforming network. The antenna is mounted on a rotatable platform under mechanical steering and motion control for aiming the antenna in the azimuth direction. Such approach significantly reduces the complexity and increases the reliability of the mechanical design. The antenna height is compatible to the two-dimensional electronic steering phased-array antenna. Additionally, the number of the electronic processing elements required is considerably reduced from that of the conventional two-dimensional electronic steering phased-array antenna, thereby allowing for low cost and large volume commercial production. 
   The present invention provides electronically generated left, right, up, and down beams for focusing the antenna beam toward the satellite while the vehicle is moving. All of the beams are simultaneously available for use in the motion beam tracking. This provides much faster response and less signal degradation. 
   The waveguide couples the EM energy from all radiating elements in the waveguide axis direction and combines the energy together. It has been found that the loss through the waveguide coupling and combining is significantly lower than that using conventional approach utilizing passive microwave processing elements printed on the circuit board at the proposed operating frequency. In addition, the present invention also reduces the number of low noise amplifiers used in the antenna system because only one set of low noise amplifiers for each waveguide is used, as opposed to conventionally use of one set of low noise amplifier for each radiating element. 
   The ridged waveguide of the present invention produced a more concentrated field line near the center line of the broadwall, thereby reducing the width of the broadwall from a typical value for a conventional rectangular waveguide to about 0.398 inches at an example frequency in the direction of broadcast satellite range of about 12.2 GHz to about 12.7 GHz. 
   The invention will be more fully described by reference to the following drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a schematic diagram of an antenna system including a mobile platform in accordance with the teachings of the present invention. 
       FIG. 2A  is a schematic diagram of an embodiment of a waveguide antenna of the present invention. 
       FIG. 2B  is a schematic diagram of a waveguide body decomposition of the waveguide shown in  FIG. 2A . 
       FIG. 2C  is a schematic diagram of the waveguide shown in  FIG. 2A . 
       FIG. 2D  is an alternate embodiment of the waveguide shown in  FIG. 2A . 
       FIG. 3  is a schematic diagram of an embodiment of a ridged waveguide. 
       FIG. 4A  is a schematic diagram of an embodiment of a L-shaped waveguide. 
       FIG. 4B  is a schematic diagram of a waveguide in decomposition of the waveguide shown in  FIG. 4A . 
       FIG. 4C  is schematic diagram of use of a dielectric material with a ridged waveguide. 
       FIG. 4D  is a schematic diagram of use of a dielectric material with a L-shaped waveguide. 
       FIG. 4E  is a schematic diagram of a waveguide antenna including the waveguide of  FIG. 4A . 
       FIG. 4F  is a schematic diagram of a waveguide antenna in decomposition including the waveguide of  FIG. 4A . 
       FIG. 5A  is a schematic diagram of an embodiment of a waveguide probe for use with the ridged waveguide. 
       FIG. 5B  is a schematic diagram of an embodiment of a waveguide probe assembled for use with the ridged waveguide. 
       FIG. 6A  is a schematic diagram of an embodiment of a waveguide probe for use with the inverted L-shaped waveguide. 
       FIG. 6B  is a decomposition of the inverted L-shaped bend and probe. 
       FIG. 7  is a schematic diagram of an embodiment of a beam forming network. 
       FIG. 8  is a schematic diagram of an embodiment of an adaptive beam-tracking system. 
       FIG. 9A  is a graph of an inferometer antenna pattern of the up and down beams at a center elevation angle at 65 degrees. 
       FIG. 9B  is a graph of an inferometer antenna pattern of the up and down beams at a center elevation angle at 35 degrees. 
       FIG. 10  is a schematic diagram of an embodiment of an adaptive beam forming system. 
   

   DETAILED DESCRIPTION 
   Reference will now be made in greater detail to a preferred embodiment of the invention, an example of which is illustrated in the accompanying drawings. Wherever possible, the same reference numerals will be used throughout the drawings and the description to refer to the same or like parts. 
     FIG. 1  is a schematic diagram of antenna system  10  in accordance with the teachings of the present invention. Waveguide antenna  12  comprises an antenna array formed of a plurality of waveguides  14  positioned parallel to each other on horizontal platform  13 . Horizontal platform  13  is rotatable under mechanical steering and motion control for aiming the antenna in the azimuth direction. 
   Waveguide axis  15  is in a direction perpendicular to the antenna aiming. Radiating surface  16  is the broad side facing the zenith direction. Radiating surface  16  of the waveguide antenna  12  includes a plurality of radiating elements  18  distributed at uniform spacing along waveguide axis  15 . Radiating element  18  provides coupling of electromagnetic (EM) energy between waveguide  14  and the free space. For example, radiating elements  18  can be X-shaped cross slots. Waveguide  14  couples the EM energy from all radiating elements  18  in the waveguide axis direction and combines the energy together. 
   In one embodiment, waveguide  14  is formed of a ridged waveguide, as shown in  FIGS. 2A–D and 3 . Walls  19  have a narrow width W 1 . For example, walls  19  can have a width of about 0.08 to about 0.12 inches. Bottom  20  includes width W 2  typically wider than width W 1 . For example, width W 2  can be in the range of about 0.450 to about 0.470 inches. Bottom  20  is coupled to bottom portion  22  of walls  19  or bottom  20  can be integral with bottom portion  22  of walls  19 . Ridge section  21  is positioned longitudinally between walls  19 . For example, ridge section  21  can have a rectangular or square configuration. Ridge section  21  has a height H 1  which is smaller than height H 2  of walls  19 . For example, ridge section can have a height H 1  in the range of about 0.18 to about 0.33 inches and walls  19  can have a height H 2  in the range of about 0.2 to about 0.35 inches. Radiating surface  16  is coupled or integral with top portion  23  of walls  19 . Radiating surface  16  is in the range of about 0.02 inch to about 0.03 inches or about 0.03 inches thick. 
   Radiating elements  18  can be positioned along the direction of waveguide axis  15  with the phase centers of the cross slots of radiating elements  18  positioned along a straight line along waveguide axis  15  and in between a center line of waveguide ridge section  21  and one of walls  19 . In one embodiment, radiating elements  18  can be placed about half a waveguide wavelength apart. For example, the length of radiating elements  18  can be about 0.3 inches to about 0.5 wavelength or about 0.4 inches to about 0.5 inches at an operating frequency of a direct broadcast signal of about 12.2 GHz to about 12.7 GHz. Radiating elements  18  can be spaced, for example, about 0.5 inches to about 1.0 inches or about 0.9 inches apart. Radiating element  18  provides circular polarization at any transverse position. For example, the crossing angle of the two slots of the cross slot of radiating element  18  can be 60 degrees to about 90 degrees. Accordingly, the present invention allows broader freedom in cross slot design thereby providing a modified shape of a three dimensional pattern produced by the cross slot radiating element. 
   A typical requirement to operate such mobile antenna in the Continental United States (Conus), is that the antenna beam is steered from about 25 degrees to about 65 degrees in elevation. It has been found that in order to achieve high antenna gain and low axial ratio in such an operating range, the antenna gain is optimized toward about 40 degrees to about 45 degrees in elevation. This can be achieved by offsetting radiating element  18  from the center of waveguide axis  15  toward one of walls  19 . The gain and axial ratio is optimized by moving the cross slot of radiating element  18  toward wall  19 . The offset creates circular polarization and also tilts the antenna beam toward the lower elevation instead of the zenith direction. When the edge of the cross slots of radiating element  18  reaches wall  19 , the highest possible elevation with good axial ratio can be achieved is determined. This provides an elevation operating range of about 25 degrees to about 55 degrees. 
   In one embodiment, one or more waveguides  14  are formed from a metal, such as aluminum stock for forming walls  19  and bottom  20  including ridge portion  21 . Radiating surface  16  is also formed of a metal, such as aluminum stock. Radiating surface  16  is attached to waveguides  14  by a dip brazing process or using a series of mounting elements, such as screws, bolts, adhesives, and laser weldments, along walls  19  of waveguide  14  to provide proper electric conductivity along the joint between radiating surface  16  and waveguides  14 . It will be appreciated that alternative methods can be used for coupling radiating surface  16  to waveguides  14 ,  40 . 
   An alternative construction is a metalized-surface plastic construction. Walls  19  and radiating surface  16  can be molded in a top piece of plastic having engaging hooks  24  along bottom portion  22  of walls  19 . Bottom  20  of waveguide  14 , including ridge section  21 , is molded as a second piece of plastic. Both the top and the bottom pieces are metalized, through a metal vapor deposit process or other processes known in the art. The top and bottom pieces can be snapped together through engaging hooks  24 , which also inserts pressure in the joint between radiating surface  16  and walls  19  of waveguides  14 , to ensure proper conductivity between the two pieces. This embodiment is suitable for low cost, mass production. 
   An antenna probe  25  is located on ends  27 ,  28  of the waveguide  14 , as shown in  FIG. 2 . Antenna probe  25  located on end  27  is used to couple a left-hand polarization signal from waveguide  14  to beam forming network  30 . Antenna probe  25  located on end  28  is used to couple a right-hand circular polarization signal from waveguide  14  to beam forming network  32 . Beam forming networks  30 ,  32  provide low noise amplification of the signal and apply progressively phase shifts to the signals from different waveguides  14  to compensate for progressive signal propagation delays before the signals from different waveguides  14  are combined. By changing the amount of the progressive phase shift, the beam can be steered to different elevation directions. 
     FIGS. 4A–B  and  FIGS. 4D–E  illustrate an alternative waveguide structure. Waveguide  40  comprises an inverted “L” shape. Wall  42  extends vertically downward from top surface  44  of waveguide  40 . For example, wall  42  can have a height H 3  in the range of about 0.3 to about 0.4 inches. The opposite wall  45  extends vertically downward from top surface  44 . For example, wall  45  can have a height H 4  in the range of about 0.05 to about 0.15 inches. The width of two walls  42  and  45  is in the range of about 0.04 to 0.12 inches. The width W 4  of the ridge portion  46  is in the range of about 0.06 to about 1.0 inches. Top surface  44  forms radiating surface  16 . A plurality of radiating elements  18  are formed in top surface  44 . Radiating elements  18  similar to those described above for waveguide  14  can be used in this embodiment. It will be appreciated that waveguide  40  can be used in all aspects of the present invention such as illustrated in the configuration of  FIG. 1 , in place of waveguide  14 . The ridged waveguide in  FIG. 3  is one embodiment of the inverted “L” shape in which H 3  is equal to H 4 . 
   The width W 3  of top surface  44  of the inverted “L” is small compared to the width of a conventional rectangular waveguide for the microwave frequency of interest to allow adjacent slotted waveguides to be close enough to eliminate grating lobes which would otherwise come into real space when the beam is scanned. For example, the width W 3  of top surface  44  can be in the range of about 0.4 to about 0.5 inches. Accordingly, waveguide  40  has a nominal internal width of about 0.32 to about 0.42 inches or about 0.35 to 0.40 freespace wavelengths facing the beam direction buried behind the face of waveguide  40 . Height H 3 , H 4 , and width W 3 , W 4  can be adjusted to slow the phase velocity in waveguide  40 . Accordingly, radiating elements  18  can be placed one waveguide wavelength apart and yet be close enough to each other to prevent grating lobes in the unscanned planes. Different variations of the L-shape waveguide  40  can be used to achieve the same radiation characteristics. Depth D 1  of ridge portion  46  can be adjusted to reduce the width W 3 . 
   Wall  42  as the vertical portion of the inverted “L” functions as a component of the waveguide width, thus enabling wave propagation similar to a conventional rectangular waveguide of a width approximately equal to the sum of wall  42  and top surface  44  of the “L”. The electromagnetic fields inside the “L” shaped waveguide  40  have a configuration which is similar to a simple dominant mode TE 1,0  rectangular waveguide. In  FIG. 2 , the electric field is forced to be zero by wall  19  on the right side. The currents in that narrow wall are vertical and give rise to a magnetic field (H-field) parallel to the axis of the waveguide. At locations to the left of that narrow, the H-field gradually becomes transverse to waveguide axis. Crossed slots or radiating elements  18  located at the proper position are then excited by the same magnitude of H LONGITUDINAL  and H TRANSVERSE  and circular polarization is achieved because the two magnetic field components are in time quadrature. 
   The use of inverted L-shape waveguide  40  allows radiating elements  18  to be more freely positioned on radiating surface  16  of waveguide  40  such that a high elevation beam with good gain and axial ratio can be achieved. The radiating element  18  position can be adjusted by adjusting height H 3 , H 4 . In contrast, the achievable antenna property (gain and axial ratio) of the ridged waveguide at high elevation angle can not be moved beyond the edge of the waveguide wall  19 , limiting the achievable antenna property at high elevation angle. 
   In one embodiment, one or more waveguides  40  are formed from a metal, such as aluminum stock for forming walls  42  and walls  45 . Radiating surface  16  including top surface  44  is also formed of a metal, such as aluminum stock. Radiating surface  16  is attached to wall  42  and wall  45  by a dip brazing process or using a series of mounting elements, such as screws, bolts, adhesives and (laser) weldments, along radiating surface  16  of waveguide  40  to provide proper electric conductivity along the joint between radiating surface  16  and waveguides  40 . It will be appreciated that alternative methods can be used for coupling radiating surface  16  to waveguides  40 . 
   An alternative construction is a metalized-surface plastic construction. Walls  42  can be molded in a top piece of plastic having engaging hooks along top portion  48  of walls  42 . Radiating surface  16 , including ridge portion  46 , is molded as a second piece of plastic. Both the top and the bottom pieces are metalized, through a metal vapor deposit process or other processes known in the art. The top and bottom pieces can be snapped together through the engaging hooks, which also inserts pressure in the joint between radiating surface  16  and wall  42  of waveguides  40 , to ensure proper conductivity between the two pieces. This embodiment is suitable for low cost, mass production. 
   Another approach to achieve high gain, low grating lobe, and good axial ratio is to employ low loss dielectric-loaded waveguide as shown in  FIG. 4C  and  FIG. 4D . The dielectric-loaded waveguide employs a low loss dielectric material to fill in the entire interior  53  of the waveguides  14 , as shown in  FIG. 4C . A dielectric material can be used to fill interior  52  of waveguide  40 , as shown in  FIG. 4D . All waveguide walls and radiating surface are formed by metal coating the dielectric material  14 . The cross-slot radiating elements  18  on the top radiating surface should be left uncoated such that the dielectric material is exposed to air in that portion. The gap between two adjacent waveguide should also be filled with metal or other conducting material. The wavelength within the dielectric material is inversely proportional to the square of the dielectric constant of the dielectric material. The use of dielectric material allows the wavelength within the waveguide to be significantly reduced, thereby suppressing the grating lobes and increasing the antenna gain. A suitable dietectric material  50  is C-Stock from Cuming Microwave and Eccostock HT003 from Emerson Cuming. 
   Referring to  FIGS. 5A–B , an embodiment of antenna probe  25  is shown. Antenna probe  25  is used for coupling electromagnetic energy between waveguide  14  and an active beam forming circuit board. Waveguide  14  includes waveguide bend  62  to rotate the feed end of waveguide  14  downward. For example, waveguide bend  62  can be about 90 degrees. Waveguide bend  62  also reverses the orientation of ridge section  21  within waveguide  14 . Antenna probe  25  is printed onto surface  63  of beam forming network printed circuit board  64 . For example, beam forming network printed circuit board  64  can be a two layered printed circuit board (PCB). Antenna probe  25  is formed as an extension of the microstrip  65 . Antenna probe  25  can have a termination  66  having a larger dimension than microstrip  65 . For example, termination  66  can be rectangular. Termination  66  is attached by microstrip  65  to ridge section  21  at lower end  67  of waveguide bend  62 . Cavity  68  under antenna probe  25  terminates waveguide bend  62 . For example, cavity  68  can have a depth of about a quarter wavelength. Through holes  69  connect to microstrip  65 . 
   Corresponding to the position of waveguide wall  19 , a grounded strip, such as formed of copper, containing a series of ground vias (not shown) forms the continuation of the waveguide wall  19 . An active low noise amplifier can follow antenna probe  25  on microwave beam forming network printed circuit board  64  to amplify the signal. The probe shown in  FIG. 5  has been analyzed using the Ansoft&#39;s EM simulation CAD tool called High Frequency Structure Simulator HFSS. It was demonstrated that less than about 0.2 dB loss can be achieved using this probe implementation. Antenna probe  25  has low loss and is easy to manufacture. The employment of the 90 degree bend allows the antenna probe is be realized as part of the PCB. Accordingly, no additional attachment mechanism is required. This is advantageous to the ease of manufacturing and reliable performance. 
     FIGS. 6A–B  illustrate an embodiment of an antenna probe which can be used with the inverted L-shaped waveguide. Waveguide  40  includes waveguide bend  72  to bring top surface  44  and ridged portion  48  of waveguide  40  downward and to a microstrip line transition. Waveguide bend  72  also converts the inverted L-shaped ridge waveguide to a symmetric ridge waveguide. For example, bend  72  can be about 90 degrees. Antenna probe  25  comprises microstrip portion  74  printed onto one side of a microwave beam forming network printed circuit board  64 . Waveguide  40  is press fit onto the microstrip portion  74  through a section of conducting block  75  and termination  76  to form the waveguide to microstrip line transition. For example, termination  76  can be rectangular or square  42 . Wall  19  is connected to the ground plane of the microstrip portion  74  through via holes  77  show in  FIG. 6 . The ground plane at the bottom of the PCB  64  terminates the waveguide. The probe implementation shown in  FIG. 6  has been analyzed by using the Ansoft&#39;s EM simulation CAD tool called High Frequency Structure Simulator HFSS. It was demonstrated that less than about 0.2 dB loss can be achieved using this probe implementation. This antenna probe waveguide termination design offers the same advantages of ease of manufacturing, low loss, and reliable performance as that in  FIG. 5 . 
   An embodiment antenna beam forming networks  30 ,  32  is shown in  FIG. 7 . Beam forming networks  30 ,  32  comprises antenna probe  25 , low noise amplifier  80 , bandpass filters  81 ,  82 , downconverter  88 , phase shift elements  86 ,  87 , and combiners  84 . Low noise amplifier  80  amplifies the received signal and bandpass filters  81 ,  82  remove the adjacent band interference and noise for each waveguide  14 ,  40  which is passed to LPF  83 . Combining network  84  combines the signal from all waveguides  14 ,  40  after the phase of the received signals from each waveguide  14  is adjusted by phase shift elements such that the signals are combined in phase. Series delay lines  86  feed local oscillator (LO) signal  87  into downconverters  88 . Series delay lines  86  can be used to generate a progressive phase shift in the local oscillator signal used in the downconverter  88  for each waveguide signal such that the signals at the output of the downconverters  88  are in phase, as described in U.S. patent application Ser. No. 10/287,370 and application Ser. No. 10/287,371, hereby incorporated in their entirety by reference into this application. Accordingly, combiners  84  add up all the signals in phase. This is the received signal which is passed to the receiver demodulator. By changing the LO frequency, different amounts of progressive phase shifts are generated, allowing the beam forming networks  30 ,  32  to steer the antenna beam to different elevation directions. Once the beam is formed, the signal is passed to frequency translator  89  to convert the signal to the desired output frequency. 
   To facilitate the in-motion pointing of the antenna beam toward a satellite, the present invention provides four additional antenna beams, such as left/right and up/down beams. Left beam  91  and right beam  92  are created by using different cross slot spacing along even and odd numbers of waveguides  14 , shown in  FIG. 1 . Wider spacing allows one beam to tilt less than the other beam using the narrower slot spacing or pitch, as shown in  FIG. 1 . Combining an odd waveguide  14  in adaptive beam forming module  90   a  creates left beam  91  and combining an even waveguide  14  in module  90   b  creates right beam  92  or vise versa depending on if a wider or narrower slot spacing is used on an odd or even waveguide, as shown in  FIG. 7 . 
   Referring to  FIG. 7 , the phase center of the beam created by the first half of waveguides  14  is at a significantly larger distance (multiple waveguide width) from the phase center of the beam created by the second half group of waveguides  14 . The distance between the phase centers allow the interferometer antenna pattern as shown in  FIGS. 9A–9B  to be created. As shown in  FIG. 7 , the combining network provides two outputs which sums up the signals from first half (1, 2, . . . 16) of the waveguides and those from the second half (17, 18, . . . 32) of the waveguides. Up beam  93  is formed by combining a 90 degree phase shifted of the first half of waveguides  14  and a second half of waveguides  14 . Down beam  94  is formed by the combining the first half of waveguides  14  and the 90 degree phase shifted of the second half of waveguides  14 . 
   In  FIG. 9A , the up beam pattern and the SUM beam pattern are shown. The SUM beam pattern points to a 65 degree elevation angle in  FIG. 9A  and the up beam points to slightly higher elevation angle by approximately 2 degrees. In  FIG. 9B , the SUM beam points to a 30 degree elevation angle and the up beam points to approximately 33 degrees. Similar pattern for down beam can be generated with down beam points approximately 2 to 3 degrees below the SUM beam. In the preferred embodiment, the 90 degree phase shifter is used to generate the up and down beam for ease of implementation. Alternatively, phase shifters with other angles can be used to create similar up and down beams with greater or smaller angle separation from the SUM beam. 
   Sum beam  98 , left beam  91 , right beam  92 , up beam  94 , and down beam  95  in mux  97 , are shown in  FIG. 7 . Satellite in-motion tracking can be accomplished by monitoring the signal powers of left beam  91 , right beam  92 , up beam  93 , and down beam  94  with power detector  99 . Left beam  91  and right beam  92  are compared against each other and sum beam  98  to obtain information regarding the antenna pointing error in the azimuth direction. Up beam  93  and down beam  94  are compared against each other and sum beam  98  to obtain information regarding the antenna pointing error in the elevation direction. The azimuth error is used to adjust the azimuth motor to dynamically move antenna platform  13 , as shown in  FIG. 1 , and the elevation error is used to adjust the electronic beam steering networks  30 ,  32  to move the beam in the elevation direction to focus the beam to the satellite during in-motion tracking of the satellite. Accordingly, the present implementation of the left/right/up/down beams allows the antenna to track the satellite during vehicle motion. The use of the four antenna beams allows the in-motion tracking to respond significantly faster than conventional systems. The antenna in-motion tracking can therefore be accomplished without or with a minimum number of motion sensors, thereby, reducing the overall cost of the system. 
   In another embodiment, in-motion antenna tracking can be used in antenna system  10 . An adaptive beam forming processing as shown in  FIG. 8  is employed in the in-motion antenna tracking system to automatically track the beam in elevation direction through the beam forming network. The adaptive beam forming processing is based on the principle of a correlating signal to derive a set of antenna weights to optimize the combined signal-to-noise ratio. By applying such operation to the output signal of each waveguide  14 , a set of antenna weights can be generated to automatically optimize the output signal-to-noise ratio. This is equivalent to precisely pointing the antenna beam to the satellite direction. The (pre-detection) signal-to-noise ratio of the output of individual waveguide is typically quite low (close to 0 dB) to typical satellite signal applications. For example, the correlation is done by multiplying two signals and then integrating (or equivalently, low pass filtering of) the output of the multiplier. The time of integration (or the bandwidth of the integration) determines the post-detection signal-to-noise ratio. Integration time of 100 uS to 1 mS can bring the post-detection signal-to-noise ratio to more than  10  dB, thereby enabling accurately determination of the antenna weight used for combining. The adaptive beam forming processing can be based on the principle of Maximum Ratio Combining (MRC), Constant Modulus Algorithm (CMA), Multiple Signal Classifications (MUSIC), or various other principles to maximize the signal-to-noise ratio. Adaptive signal processing is applied to the elevation angle tracking for antenna system  10 . In a two dimensional phased-array antenna, the adaptive signal processing technique can be applied to track the signal in both elevation and azimuth direction. 
   An embodiment of the adaptive beam tracking system  100  based on MRC is illustrated in  FIG. 10 . The signals from a plurality of waveguides  14 ,  40  are input into the beam forming processing. It will be appreciated that various numbers of antennas and processing elements could be used in accordance with the teachings of the present invention. Modulators  102   a–d  apply determined antenna weights  103  to the signal. Modulators  102   a–d  are controlled by the antenna weight to generate the desired phase shift and gain scaling for the signals. The outputs of modulators  102   a–d  are combined in summer  104  to generate combined (beam formed) output signal  106 . 
   The antenna weight is computed by downconverting the input signals and the combined signal to baseband. In one embodiment, a direct down-conversion processing is employed in which the LO frequency is the same as the input signal frequency. The signal is thereby converted to the baseband. The output of the downconverter is first filtered to extract the signal in the desired frequency band. The signals from plurality of waveguides  14 ,  40  are downconverted in respective downconverters  110   a–d.  Each of downconverters  110   a–d  multiplies the signal from a different waveguide  14  by a local oscillator in-phase signal (LOI) and a local oscillator quadrature phase signal (LOQ). The resultant signals are applied to respective low-pass filters (LPF)  112   a,    112   b  in a baseband automatic gain control (AGC) loop  116  that normalizes the signal level before the MRC algorithm. AGC loop  116  provides a consistent performance at different input signal levels. Variable gain amplifiers  118   a,    118   b  are applied to the respective outputs of LPF  112   a,    112   b  and MRC beamforming module  120 . At the output of the variable gain amplifiers  118   a,    118   b,  power detectors  117  are applied to sum the signal power of all antennas and compare the signal power to a threshold value. The difference between the signal power of all antennas and the threshold value can be integrated to maintain the signal level after AGC loop  116  at the same level and can be used to adjust the gain of variable gain amplifiers  118   a,    118   b.  Accordingly, in this implementation, the MRC algorithm is able to work at different input signal levels. 
   MRC beamforming module  120  performs real time adaptive signal processing to obtain the maximum signal-to-noise ratio. In an implementation of MRC beamforming module  120  the antenna weights are used to align the phases of the four antenna signals received from waveguides  14  and also scale the signal in proportion to the square-root of the signal-to-noise ratio in each individual channel. In one implementation, the signal envelope is used as an approximation to scale the signal in proportion to the square-root of the signal-to-noise ratio in each individual channel. 
   MRC beamforming module  120  can employ a Cartesian feedback loop. MRC beamforming module  120  provides baseband processing which performs complex conjugate multiplication of the output of a baseband I and Q channel filter with a baseband reference I and Q channel as follows:
 
 I   — ERROR i   =I   i   *I   s   +Q   i   *Q   s 
 
 Q   — ERROR i   =I   i   *Q   s   −Q   i   *I   s 
 
   The resultant signal (I — ERROR i , Q — ERROR i ) at the output of MRC beamforming module  120  is a complex signal with phase equal to the difference of the reference complex signal and the individual signal and an envelope proportional to the envelope of the individual signal. Signal I — ERROR is applied to integrator  122   a  and signal Q — ERROR is applied to integrator  122   b.  The output of the LPFs  122   a,    122   b  is antenna weight  103  (IWi, QWi, i=1, 2, 3, . . . ). The antenna weight normalization computes the summation of all the antenna weight and normalizes the summation to a constant through the use of the feedback operation. 
   Combined signal  106  is applied to downconverter  128  and is multiplied by LOI and LOQ. The resultant signals are applied to low-pass filters (LPF)  130   a,    130   b.  The outputs from the low-pass filters (LPF)  130   a,    130   b  are amplified with quadrature phase signal amplifiers  131   a,    131   b  and applied to antenna weight magnitude normalization module  132 . 
   Antenna weight magnitude control loop  132  monitors the power in the combined signal. If the magnitude of the weight is small, the power of the combined signal is small. Alternatively, if the magnitude of the weight is large, the power of the combined signal is large. A power detector can be used in the antenna weight magnitude control loop  132  to compare the power of combined signal  106  with a threshold level. The difference between the power of combined signal  106  and the threshold level is filtered such as with a low-pass filter (LPF). The filtered output can be fed forward to the variable gain amplifiers to adjust the magnitude of the combined signal. A higher gain in the variable gain amplifiers produces a larger antenna weight and a lower gain in the variable amplifiers produces a smaller antenna weight. By varying the gain of the variable gain amplifiers in the baseband SUM channel signal paths, the magnitude of the antenna weight is adjusted to a proper level to keep the output signal power in a small range. 
   Output from antenna weight magnitude normalizing module  132  is amplified with quadrature phase signal amplifiers  134   a,    134   b  and is applied to MRC beamforming module  120  to be used for updating antenna weight  103 , as described above. 
   An advantage of the adaptive beam forming processing of the present invention is a fast response and reliable tracking in the elevation beam. This is achieved via the processing on the phase of the signal directly instead of processing on the signal power as in the conventional elevation tracking system. Generally, the adaptive processing of the present invention achieves fast and reliable performance in a much lower signal-to-noise ratio. Additionally, the adaptive processing as illustrated in  FIG. 10  is amendable to integrated circuit processing, thereby, reducing the overall cost of the system. Another advantage of present invention is that the overall tracking can be greatly simplified because the system now only needs to monitor the power of left and right beam and command the motor to move the antenna to track in azimuth direction. Accordingly, no motion sensors are used. 
   It is to be understood that the above-described embodiments are illustrative of only a few of the many possible specific embodiments, which can represent applications of the principles of the invention. Numerous and varied other arrangements can be readily devised in accordance with these principles by those skilled in the art without departing from the spirit and scope of the invention.