Abstract:
A data comparator that operates on an input voltage signal and a reference voltage signal is disclosed. Internally, the comparator includes replicated circuitry to produce differential gain. Each set of replicated circuitry includes two gain stages for high amplification, high sampling rate, and for reducing kickback noise at the input voltage signal and the reference voltage signal. The comparator may further include self-biased CMOS inverters for cancellation of input offset error and a rail-to-rail regenerative output latch. The circuit can also include a comparator bias circuit that can improve the speed of the auto-zero operation.

Description:
RELATED APPLICATIONS 
     Priority is claimed to the following Patent Applications: 
     U.S. Provisional Patent Application No. 60/311,696, entitled “Fast Hybrid Data Comparator,” filed on Aug. 10, 2001; 
     U.S. Provisional Patent Application No. 60/311,548, entitled “Hybrid Data Comparator,” filed on Aug. 10, 2001; and 
     U.S. Provisional Patent Application No. 60/311,556, entitled “Starved Hybrid Data Comparator,” filed on Aug. 10, 2001; 
     The entirety of these Patent Applications is expressly incorporated herein by reference. 
    
    
     BACKGROUND 
     1. Technical Field 
     The field of the invention is related to data comparators and more particularly to chopper-topology data comparators. 
     2. Description of Related Art 
     Data comparators are commonly used when an analog signal (e.g., one that can assume continuously variable voltage and/or current characteristics versus time) must be translated to a digital format. The translation can be performed by an analog-to-digital converter, or “ADC”, which is comprised of a number of data comparators. The act of translation is referred to as sampling or digitization. 
     Applications requiring digitization include, but are not limited to, the digital recording of audio and/or video images, the digital processing of audio and/or video images for transmission or reception, as part of a transducer for processing of non-electrical signals such as mechanical or optical or magnetic sensors, etc. A variety of low-power electronic devices such as mobile stations, personal digital assistants, and digital cameras, to name but a few, rely on digitization of analog signals. 
     The performance of ADCs may be specified in terms of several basic characteristics including, but not limited to, power dissipation, maximum sampling rate, minimum voltage supply, noise immunity, and complexity. The data comparator drives the fundamental limits of each of these characteristics. Required specifications vary considerably among the applications of ADCs, but many require low power (microwatts/comparator), sample rates over 50 MSPS (50×10 6  samples/second), low voltage supply (down to 1V or less), and excellent noise immunity (typically produced by differential signaling). 
     A known topology that facilitates good characteristic specifications is the chopper topology. The topology features a gain stage, a continual auto-cancellation of offset voltage errors that accrue at the input of the gain stage, relatively low power consumption, high sampling rates, and low voltage supply operation. Several enhancements to the basic chopper topology can further improve circuit characteristics and comprise the prior art. However, prior art chopper-topology comparators typically compromise one or more of the specifications to favor the remainder. 
     For example, several data comparator topologies require an explicit output latch as an auxiliary circuit. Others require a relatively large number of passive charge storage elements (capacitors), and some have complex clock sequences. Still others, although utilizing differential signaling, make use of only half the differential voltage between the input voltage signal and the reference voltage signal. 
     Some data comparators employ only a single gain stage, which reduces the differential gain, requires a higher-gain output latch, and leaves the input voltage signal and reference voltage signal susceptible to kickback noise generated by the output latch. Most data comparators also lack a mechanism to prevent unnecessary, excessive current draw by the inverter amplifiers in the latched state. 
     While typical chopper-topology comparators can satisfy various performance criteria in piecemeal fashion, their ability to do so is degraded as voltage supplies are reduced below 1.5 V due to the aforementioned deficiencies. Thus, a data comparator that offers high performance in all performance categories while operating at low input voltages is desired. 
     SUMMARY 
     In a first principal aspect, a method of measuring an input voltage using a first half-cell comparator and a second half-cell comparator is disclosed, wherein each half-cell comparator includes a first amplification stage, a second amplification stage, at least one first coupling component for coupling voltage to the input of the first amplification stage, and at least one second coupling component for coupling an output of the first amplification stage to an input of the second amplification stage. The method may include applying, during an auto-zero time period, the input voltage to the first coupling component of the first half-cell comparator, and also applying a reference voltage to the first coupling component of the second half-cell comparator. 
     The method may also include applying, during a sample time period, the reference voltage to the first coupling component of the first half-cell comparator, and also applying the input voltage to the first coupling component of the second half-cell comparator during the sample time period. 
     During a third time period, the latching period, the output of the second amplification stage of the first half-cell comparator may be coupled to the input of the second amplification stage of the second half-cell comparator; also during the third time period, the output of the second amplification stage of the second half-cell comparator may be coupled to the input of the second amplification stage of the first half-cell comparator. 
     In a second principal aspect, a method of measuring an input voltage using a first half-cell comparator and a second half-cell comparator is disclosed. The method is similar to the method discussed above, but further includes receiving at a control circuit, during a third time period, complementary half-cell outputs from the first and second half-cell comparators, wherein one output is substantially equal to a maximum supply voltage and wherein the complementary output is substantially equal to a minimum supply voltage. 
     Also during the third time period, the method includes outputting from the control circuit a maximum and a minimum control voltage, wherein the maximum and the minimum control voltage are substantially the same regardless of which half-cell comparator output is substantially equal to the maximum supply voltage and regardless of which half-cell comparator output is substantially equal to the minimum supply voltage. 
     The method further includes receiving, during the third time period, the maximum and minimum control voltages at the first amplification stage of each half-cell comparator, and disabling the first amplification stage of each half-cell comparator in response to the control voltages. 
     In a third principal aspect, a voltage measuring circuit comprising a first half-cell comparator and a second half-cell comparator is disclosed. Each half-cell comparator can include a first amplification stage, at least one first coupling component (such as a capacitor) for coupling voltage to the input of the first amplification stage. The circuit may also include a first pair of complementary clocked analog switches for applying either a reference voltage or an input voltage to each first coupling component, the first pair of complementary clocked analog switches connected so that when the input voltage is applied to the first half-cell, the reference voltage is applied to the second half-cell, and when the reference voltage is applied to the first half-cell, the input voltage is applied to the second half-cell. 
     The circuit may further include a second amplification stage and at least one second coupling component for coupling the output of the first amplification stage to the input of the second amplification stage. The output of the second amplification stage can be latched by a cross-coupling clocked analog switch that connects the output of the second amplification stage of one half-cell to the input of the second amplification stage of the other half-cell. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The above and other features of the present invention will become more apparent from the following detailed description taken with the accompanying drawings in which: 
     FIG. 1A is a schematic view of a comparator in accordance with an exemplary embodiment of the invention; 
     FIG. 1B is a schematic view of a fast comparator in accordance with an alternative exemplary embodiment of the invention; 
     FIG. 1C is a schematic view of a low-power comparator in accordance with another alternative exemplary embodiment of the invention; 
     FIG. 2A illustrates a switch level representation of an inverter that may be used in accordance with the invention; 
     FIG. 2B illustrates an inverter amplifier that may be used in accordance with the invention; 
     FIG. 2C illustrates an inverter amplifier with control terminals that may be used in accordance with a low-power embodiment of the invention; 
     FIG. 3A illustrates an analog switch that may be used in accordance with the invention; 
     FIG. 3B illustrates an alternative analog switch that may be used in accordance with the invention; 
     FIG. 3C illustrates another alternative analog switch that may be used in accordance with the invention; 
     FIG. 4 shows the waveforms of clock signals applicable to control the comparator of the exemplary embodiment; 
     FIGS. 5A-5E illustrate NMOS implementations of a bias circuit that may be used in accordance with the invention; 
     FIGS. 5F-5J illustrate PMOS implementations of a bias circuit that may be used in accordance with the invention; and 
     FIG. 6 illustrates a control circuit that may be used in accordance with a low-power embodiment of the invention. 
    
    
     DETAILED DESCRIPTION 
     FIG. 1A is a schematic diagram of a chopper-topology comparator in accordance with an exemplary embodiment of the invention. It is contemplated that the comparator would be fabricated in quantity, along with additional circuitry such as switches and other components, to implement a complete analog-to-digital comparator using existing integrated circuit (IC) fabrication techniques and materials (e.g. CMOS, BiCMOS, silicon, gallium arsenide, germanium, etc.). A typical use for a number of comparators as described here could include, for example, 8-bit, 12-bit and 16-bit, 300MS/s flash ADCs, where each comparator produces thermometer code output that can be used to create a binary coded output of the desired number of bits. 
     The comparator may be designed to operate from a range of supply voltages. The upper limit of supply voltage is determined by reliability constraints such as dielectric breakdown, punch-through, avalanche breakdown, and hot carrier effects. Typical values in present state-of-the-art manufacturing processes range from 3V to 5V. The lower limit of supply voltage is determined by the transistor threshold voltage (VT) such that the minimum voltage is approximately twice VT. Typical VT values in present state-of-the-art manufacturing processes are about 0.3 V to 0.6 V. Thus, exemplary embodiments of the comparator using state-of-the-art and future semiconductor manufacturing processes can achieve sampling rates of 1 to 1,000 Megasamples per second (MSPS) using a supply voltage of about 0.5V to about 5V. In operation, the comparator could consume from about 10 nanowatts to about 100 microwatts of power, although operation at other power levels is possible. 
     In an exemplary embodiment, the comparator of the present invention comprises two half-cells operated differentially. Each half-cell may include 1) an input selector, 2) first- and second-stage inverting amplifiers, and 3) cross-coupling analog switches to implement a regenerative latch. Functional equivalents to analog switches are also possible. FIG. 2A is a switch level representation of an inverter circuit that can be used in accordance with the invention. When S 1  is open and S 2  is closed, as shown, the circuit is equivalent to a high-input, low-output condition, where S 2  discharges the capacitive load C L . When S 2  is open and S 1  is closed (not shown), the circuit is equivalent to a low-input, high-output condition, since S 1  will charge C L . FIG. 2B illustrates a possible configuration of one of the inverter amplifiers that may be used in exemplary embodiments of the invention, while FIG. 2B illustrates an alternative inverter amplifier that may be used in accordance with a low-power or “starved” hybrid comparator, to be described in detail below. 
     A number of analog switches may be used in accordance with the invention, and FIGS. 3A,  3 B, and  3 C show several specific configurations of an analog switch that may be used in exemplary embodiments of the invention, although the particular configuration of any analog switch is not necessarily critical to all embodiments of the invention. Switches used in accordance with the invention can be designed (i.e., switch width can be chosen) to operate in the linear region for fast recovery. Besides designing for fast recovery, the complementary configuration of FIG. 3A can be utilized to cancel clock feedthrough currents that contribute to systematic error in the DC quiescent point. For the complementary configuration, low resistance is achieved by ensuring the transistors operate in the linear region. A switch may therefore be modeled as a single NMOS with a transconductance equal to the combination of both transistors. As an additional design consideration, a wide switch results in a low drain-source voltage, but the additional capacitance can result in a slow, underdamped response, whereas a narrow switch has a fast underdamped response, so a narrow switch may be preferred. 
     Switches S 6  and S 8  (FIG. 1A,  1 B, and  1 C) form a complementary clocked pair of analog switches; when S 6  is closed, S 8  will be open, and vice-versa. Switches S 5  and S 7  (FIGS. 1A,  1 B, and  1 C) form a similar complementary clocked pair for the other half-cell comparator, with input voltages reversed from those of S 6  and S 8 , so that when an input voltage is applied to the first half-cell, a reference voltage is applied to the second half-cell, and when the reference voltage is applied to the first half-cell, the input voltage will be applied to the second half-cell. 
     Switches S 1 , S 2 , S 3 , and S 4  (FIGS. 1A,  1 B, and  1 C) may be referred to as clocked auto-zero analog switches; switches S 9  and S 10  (FIGS. 1A,  1 B, and  1 C) may be referred to as cross-coupling clocked analog switches; and switches S 11  and S 12  (FIG. 1B) may be referred to as clocked bias-control analog switches. 
     The clock sequencing of the comparator is illustrated in FIG.  4 . As indicated, three time periods are produced and performed repeatedly. Two clock signals, K 1  and K 2 , represent the logic defining the time periods. The logic complements, K 1   b  and K 2   b , can also be used to control the comparators. 
     Auto-Zero Period 
     In the first time period, also referred to as the auto-zero period, K 1  is asserted (at its maximum voltage) and K 2  is deasserted (at its minimum voltage). This causes analog switches S 1 , S 2 , S 3 , S 4 , S 7 , and S 8  to close and switches S 5 , S 6 , S 9 , and S 10  to open. During this period, each inverting amplifier I 1 , I 2 , I 3 , and I 4  has its input and output coupled together by the analog switches, which causes the input and output voltage to assume a potential of approximately one-half of the voltage supply due to the quiescent operating current that results from the geometry of the transistors of the inverting amplifiers when the output voltage equals the input voltage. The NMOS and PMOS transistors of each half-cell&#39;s inverters may be sized so the inverter threshold is about Vdd/2, which implies β n ≈β p  and V T,n ≈V T,p . 
     Both transistors of an inverter are assumed to be operating in the saturation region, corresponding to the small-signal input and output relative to a V dd /2 reference during auto-zero. This synchronous auto-zero operation reduces or eliminates common mode noise, and also reduces or eliminates input offset voltage error that is inherent in some ADC comparators that use differential pair topologies. 
     During auto-zero, capacitor C 1  samples the potential of the reference voltage signal VREF because S 7  is closed and S 5  is open. Similarly, capacitor C 2  samples the potential of the input voltage signal VIN because S 8  is closed and S 6  is open. In a similar manner, capacitor C 3  samples the potential of the output of the inverting amplifier I 1  and capacitor C 4  samples the potential of the output of the inverting amplifier I 2 . 
     Sample Period 
     In the second time period, also referred to as the sample period, K 2  remains at minimum voltage and K 1  is deasserted (at its minimum voltage), causing analog switches S 1 , S 2 , S 3 , S 4 , S 7 , and S 8  to open, and causing switches S 5 , S 6 , S 9 , and S 10  to close. Since the inputs to the first stage of each half cell are reversed, the half cells always operate 180 degrees out of phase with each other so that one produces a positive differential signal while the other produces a negative differential signal, as follows. During the sample period, each inverting amplifier I 1 , I 2 , I 3 , and I 4  is free to operate on any signal at its input and deliver the amplified and inverted signal to its output. Also during this period, capacitor C 1  now samples the potential of the input voltage signal VIN because S 5  is closed and S 7  is open, and capacitor C 2  now samples the potential of the reference voltage signal VREF because S 6  is closed and S 8  is open. This causes a differential voltage (VIN−VREF) to be transferred to C 1  and a differential voltage (VREF−VIN) to be transferred to C 2 . The differential signals propagate through first-stage inverting amplifiers I 1  and I 2  and are transferred to capacitors C 3  and C 4 , and then through second-stage inverting amplifiers I 3  and I 4 . Since two inverting amplifiers are present in each half-cell, the half-cell input and output signals have the same polarity. At the end of the second time period, the half-cell outputs Q and Qb have amplified signals corresponding to (VIN−VREF) and (VREF−VIN), respectively. 
     In an equivalent small-signal model, each half-cell acts as a cascade of two low-pass filters (which can be modeled as a passive RC filter) and two small-signal inverting amplifiers. The “resistance” of the low-pass filters is contributed by the small-signal impedance of C 1  and C 2 , and the capacitance is due to the relatively small inverter input capacitances C i . The effect of the equivalent low-pass filter is negligible, however, for the following reason. A small-signal input of ν Δ0  to the first equivalent filter is attenuated by C/C+C i , but C is much larger than C i , so the attenuation factor is approximately 1. 
     Within each half cell, the first inverter provides a first stage small-signal gain, and the second inverter provides a second stage small-signal gain to further boost the differential gain prior to latching. If required, more stages could be used to provide additional gain. Since each half cell is fully differential, excellent power supply rejection is realized. 
     Latch Period 
     In the third time period, also referred to as the latch period, K 1  remains at minimum voltage and K 2  is asserted (at its maximum voltage), causing analog switches S 9  and S 10  to close. During this period the last inverting amplifier in each half-cell (I 3  and I 4 ) is connected in an integral positive-feedback regenerative latch via analog switches S 9  and S 10 . As in the auto-zero period, the analog switches are designed to ensure operation in the linear region. As a result of the positive feedback, the differential voltages present at outputs Q and Qb from the sample period are driven to full rail potentials (i.e., to approximately the maximum or minimum possible potential), where they remain irrespective of changes in the input voltage. During latching, one transistor in an inverter will be saturated and the other will be operating in linear mode, resulting in the final rail-to-rail outputs. The latch period concludes when K 1  is asserted (at its maximum voltage) and K 2  is deasserted (at its minimum voltage). When K 1  is asserted, switches S 7 , and S 8 , S 1 , S 2 , S 3 , and S 4  close. Since K 1   b  is deasserted, switches S 5 , S 6  will open, while K 2  causes switches S 9  and S 10  to open, and operation returns to the auto-zero period. 
     In the exemplary embodiment, the use of two gain stages in each half-cell shields the input voltage signal and the reference voltage signal from kickback noise produced by the large voltage transitions of the regenerative output latch. As the second-stage inverters (I 3  and I 4 ) latch, the first-stage inverters (I 1  and I 2 ) create virtual grounds at capacitors C 3  and C 4 , which blocks the backwards propagation of kickback noise from the outputs Q and Qb to the inputs VREF and VIN. The use of two gain stages also increases the speed of the latching operation since the voltage levels at Q and Qb begin closer to their latched values. 
     As another feature, the use of replicated input selectors using the analog switch pairs S 5 /S 7  and S 6 /S 8  permits each half-cell to operate on the full differential signal |VIN−VREF|. 
     In operation, as clock signal K 1  oscillates between its maximum and minimum voltage, the input voltage signal VIN is alternately connected to C 1  via S 5  or to C 2  via S 8 . Consequently, if C 1  and C 2  have equal capacitance then the capacitive loading of VIN is nearly constant throughout all time periods. Similarly, the reference voltage signal VREF is alternately connected to C 1  via S 7  or to C 2  via S 6 , so if C 1  and C 2  have equal capacitance then the capacitive loading of VREF is nearly constant throughout all time periods. This uniform loading contributes to improved stability of the potentials at VIN and VREF. 
     Experimental prototypes of the comparators were fabricated in a 0.5 μm, single-poly, triple-metal CMOS technology with transistor threshold voltages of V Tn =0.72V and V Tp =−0.085V. A 2.25−fF/μm 2  poly/n+ linear capacitor option was used. Two versions of the hybrid comparator were fabricated, with sampling capacitance of 400 fF and 800 fF, respectively; die areas were 414×56 μm 2  and 494×56 μm, respectively. A single version of the starved hybrid comparator, described below, was fabricated with a sampling capacitance of 400 fF; die area for this version is 492×56 μm 2 . Measured input offset in all versions is about 250 μV. Measured harmonic distortion, known to underestimate actual performance, was generally better than −30 dB. Simulations indicated successful latching to 500 MS/s, making the performance of these comparators among the highest reported in terms of sample rate, power dissipation, and low-voltage operation. 
     High-Speed Operation 
     FIG. 1B is a schematic diagram of a high-speed chopper-topology comparator in accordance with an alternative exemplary embodiment of the invention. The topology is similar to that of the basic hybrid comparator of FIG. 1A, with the addition of one or more bias circuits as shown in FIGS. 5A-5J. The bias circuit in this embodiment can shorten the auto-zero period by initiating the auto-zero of capacitors C 3  and C 4  during the latching period rather than during the subsequent auto-zero period, with a corresponding increase in sample rate. In order to take advantage of this feature, it may be desirable (but not necessary) to use asymmetric clock signals (e.g., an auto-zero period that is shorter than the other periods; other differences in timing are possible as well). During the latching period, the bias circuit is applied to C 4  and C 5  because K 2  is asserted, which closes clocked bias-control switches S 11  and S 12 . The circuits of FIGS. 5A-5J also provide a low-impedance noise discharge path that can significantly reduce the time required for decay of latch kickback noise. Faster decay of kickback noise allows for a shorter latch time period, thus, an increased maximum sample rate. Without such a low-impedance path, larger devices must be used to achieve high sampling rates, which results in a commensurate increase in power consumption. 
     K 2  or K 2   b  also turns on or “clocks” the drive transistors of the bias circuit(s) during the latch period, making it active. In the self-biased embodiments of FIGS. 5B,  5 C,  5 F,  5 G,  5 H, and  5 K, the MOSFET gate dimensions and/or resistor values are designed to ensure that the potential of the output terminal Y is approximately one-half the voltage supply when the bias circuit is active. Similarly, in the externally-biased embodiments of FIGS. 5D,  5 E,  5 I, and  5 J, the MOSFET gate dimensions and the external voltage signal VB ensure that the potential of the output side terminal Y is approximately one-half the voltage supply when the bias circuit is active. Of course, other bias voltages are possible as well, and may be desirable in specific applications. 
     The bias circuit can be isolated from the comparator by analog switches S 11  and S 12  (FIG. 1B) so it does not perturb the nodal voltages during the auto-zero or sampling period. In addition, to prevent unnecessary current draw in the bias circuit during those periods, clock signal K 2  is applied to terminal T of NMOS-input embodiments of the bias circuit (FIGS. 5A-5E) and complementary clock signal K 2   b  is applied to terminal T of PMOS-input embodiments of the bias circuit (FIGS.  5 F- 5 J), disabling the bias circuit when it is not needed. 
     Low Power Operation 
     FIG. 1C is a schematic diagram of a starved hybrid chopper-topology comparator in accordance with an alternative exemplary embodiment of the invention. In addition to the basic configuration of FIG. 1A, the low power starved comparator circuit of FIG. 1C includes a comparator control circuit as shown in FIG. 6 whose outputs, T and Tb, can be applied to the control terminals T and Tb of the inverting amplifier shown in FIG.  2 B. The two control terminals are driven when the inverter can be inactive in order to reduce or eliminate power consumption in the first amplifier stage during a particular period or periods. 
     As discussed above, during the latch period, comparator outputs Q and Qb are at or near full rail potential. Referring now to FIG. 6, regardless of whether Q is high and Qb is low or vice-versa, control circuit output T will be low and Tb will be high whenever Q and Qb are at opposite supply rails, for the following reasons. First, assume Q is high and Qb is low; Q will drive NMOS transistor N 1  “on”, so there will be a low impedance path from the drain (which is low due to Qb) to the source, which is tied to output T. Transistor N 2  will be driven “off” by Qb, and thus will not affect the voltage at T. Conversely, when Q is low and Qb is high, the source of N 2  will pull T low, and N 1  will be off. Similarly, when Q is high and Qb is low, Qb will drive PMOS transistor P 2  “on”, so there will be a low impedance path from the drain (which is high due to Q) to the source, which is tied to output Tb. Transistor P 1  will be driven “off” by Q, and thus will not affect the voltage at Tb. Conversely, when Q is low and Qb is high, the source of P 1  will pull Tb high, and P 2  will be off. The following table indicates the operation of the control circuit. 
     
       
         
               
               
               
               
               
             
           
               
                   
                   
               
               
                   
                 Q 
                 Qb 
                 T 
                 Tb 
               
               
                   
                   
               
             
             
               
                   
                 1 
                 0 
                 1 
                 0 
               
               
                   
                 0 
                 1 
                 1 
                 0 
               
               
                   
                 0 &lt; Q &lt; 1 
                 0 &lt; Qb &lt; 1 
                 bias voltage 
                 bias voltage 
               
               
                   
                   
               
             
          
         
       
     
     Referring now to FIG. 2B, it can be seen that a low input to the T control terminal will drive the lower NMOS transistor to cutoff, and the high input to the Tb control terminal will drive the upper PMOS transistor to cutoff, thus “starving” inverting amplifiers I 1  and I 2  during the latch period when they can be inactive. 
     Both the high-speed and the low-power or starved comparator configurations (or selected features of them) can be implemented at the same time, yielding very fast operation while consuming minimal power. In addition, various features described above can be used while others are left out, resulting in combinations other than those described in detail here. 
     Although several possible embodiments of an apparatus and method have been described, various changes and modifications may be made or suggested by those skilled in the art without departing from the spirit or scope of the claims that follow.