Abstract:
An electromagnetic transponder and a method for controlling by pulse trains a switch for modulating the load of this transponder in an electromagnetic field from which it extracts its power supply, the duty ratio of the pulses being controlled according to the transponder supply voltage.

Description:
PRIORITY CLAIM 
     The present application claims the benefit of French Patent Application No. 04/50950, filed May 14, 2004, which application is incorporated herein by reference in its entirety. 
     TECHNICAL FIELD 
     Embodiments of the present disclosure relate generally to the field of transponders, and more specifically to contactless smart cards and labels, intended to communicate with a read/write terminal by radio frequency data transmission. 
     Embodiments of the present disclosure more specifically apply to remotely-supplied transponders which extract the power necessary to their operation from the electromagnetic field radiated by the terminal close to which they are located. 
     BACKGROUND 
       FIG. 1  shows, in the form of blocks, an electromagnetic transponder read terminal and a transponder which communicates with this terminal. 
     On the read terminal side are generally found a series oscillating circuit  2  formed of an inductance L 1  forming an antenna in series with a capacitor C 1  coupled between an output terminal  3  of an amplifier or antenna coupler (not shown) and a reference terminal  4  (generally, the ground). Generally, a resistor (not shown) is interposed between the antenna coupler output and inductance L 1 . The antenna coupler belongs to one or several circuits  5  (LECT) for controlling the oscillating circuit and exploiting received data and comprises, among others, a modulator-demodulator and a microprocessor for processing the control and data signals. In the example of  FIG. 1 , a node  6  of interconnection between capacitor C 1  and inductance L 1  forms a terminal for sampling the data signal received from transponder  10 , intended for the demodulator of terminal  1 . 
     Circuit  5  of the terminal generally communicates with different input-output circuits (keyboard, screen, means of transmission to a central server, etc.) and/or processing circuits not shown. The circuits of read/write terminal  1  extract the power necessary to their operation from a supply circuit (not shown) coupled, for example, to the electric system or to a battery. 
     On the side of transponder  10 , an inductance L 2 , in parallel with a capacitor C 2 , forms a parallel oscillating circuit (called a resonant circuit) intended to sense the electromagnetic field generated by the series oscillating circuit (L 1 , C 1 ) of terminal  1 . The resonant circuit (L 2 , C 2 ) of transponder  10  is tuned to the resonance frequency of oscillating circuit L 1 C 1  of terminal  1 . Parallel resonant circuit L 2 C 2  is intended to capture the electromagnetic field generated by series oscillating circuit L 1 C 1  of terminal  1 . 
     Terminals  11 ,  12  of resonant circuit L 2 C 2 , corresponding to the terminals of capacitor C 2 , are coupled to two A.C. input terminals of a rectifying circuit  13  (RED) formed, for example, of a bridge (not shown) of four diodes for a fullwave rectification. It should be noted that circuit  13  may be replaced with a halfwave rectifying assembly. A capacitor Ca is coupled to the rectified output terminals  14  and  15  of circuit  13  to store the power and smooth the rectified output voltage. 
     When transponder  10  enters the electromagnetic field of terminal  1 , a high-frequency voltage is generated across resonant circuit L 2 C 2 . This voltage, rectified by circuit  13  and smoothed by capacitor Ca, becomes a supply voltage (VD) on terminal  14  intended for electronic circuits  20  of transponder  10 . Voltage VD is generally regulated by a voltage regulator (not shown). Circuits  20  (CTL) generally comprise a microprocessor, a memory, and a control circuit. For simplification, reference will be made hereafter to a single circuit  20  containing all these elements. Transponder  10  is generally synchronized by means of a dock (CLK) extracted, by a block  18 , from the high-frequency signal recovered across capacitor C 2 , before rectification. Circuit  20  contains a means for demodulating signals  17  (DMOD) coming from terminal  1  and sampled on terminal  14 . To transmit data from transponder  10  to terminal  1 , circuit  20  controls a stage of modulation (called back-modulation) of resonant circuit L 2 C 2 . This modulation stage is generally formed of an electronic switch S and of a resistor R, in series between terminals  14  and  15 . Transistor S is controlled by pulses at a so-called sub-carrier frequency (for example 424 kHz), smaller than the frequency of the excitation signal of oscillating circuit L 1 C 1  of terminal  1  (for example, 13.56 MHz). Circuit  20  provides an output signal  19  (RMOD) of pulse control of switch S. Circuit  20  and the back-modulation stage altogether form a variable load of the transponder, equivalent to a resistor added in parallel on capacitor Ca. This equivalent resistor will be designated hereafter as Req. 
     When switch S is on, the oscillating circuit of the transponder is submitted to an additional damping with respect to the load formed of circuit  20 , so that the transponder samples a greater amount of power from the high-frequency electromagnetic field. Accordingly, the power variation of transponder  10  translates, on the side of terminal  1 , as an amplitude and phase variation of the current in antenna L 1 . This variation is detected in the signal present on node  6  and intended for the demodulator of terminal  1 . The oscillating circuits of terminal  1  and of transponder  10  are generally tuned to the carrier frequency, that is, their resonance frequency is tuned to the 13.56-MHz frequency. Generally, all the electronic circuits of transponder  10  are integrated in a same chip. 
       FIGS. 2A and 2B  illustrate, in the form of timing diagrams, a conventional example of the operation of transponder  10  in a data transmission to terminal  1 .  FIG. 2A  shows an example of the shape of control signal RMOD of switch S provided by circuit  20  at a sub-carrier frequency of period (T). In this example, the data are coded by trains of four pulses. Other shapes are possible. Generally, the conduction duty ratio of switch S is set to 0.5. 
       FIG. 2B  illustrates the shape of rectified voltage VD present on terminal  14 . Each turning-on of switch S forms an additional charge sampled from the power reserve stored in capacitor Ca. Back-modulation resistor R is selected to cause a power variation of transponder  10  such that the resulting current amplitude and phase variation in antenna L 1  is exploitable by circuit  5  of terminal  1 . Generally, the internal consumption current of transponder  10  increases as soon as switch S, by its turning-on, reduces equivalent resistance Req by a ratio of from three to six. This current consumption increase causes a decrease in voltage VD. 
     Conversely, each turning-off of switch S causes an increase in voltage VD until the next turning back on of switch S. In the example of  FIG. 2B , the variation ratio of resistance Req and thus of the internal consumption current of the transponder has been selected to be close to three. Thus, voltage VD generally decreases three times faster during on periods of switch S than it increases during its off periods. This weakening of voltage VD is all the more significant as transponder  10  is loosely coupled with terminal  1 , that is, when the antennas are distant from each other (by more than sixty centimeters). As a result, the regulator is no longer able to provide a sufficient supply voltage to circuit  10 . In  FIG. 2B , the minimum threshold of voltage VD short of which regulator “uncouples” and circuit  10  is no longer powered has been designated as VDMIN. In the example of  FIG. 2B , it is assumed that voltage VD reaches minimum value VDMIN at the end of a train of four pulses. 
     A problem which is posed when the oscillating circuits of the transponder and of the terminal are distant is that the power sampled by the back-modulation circuit is such that the transponder supply voltage is decreased to the point that its internal circuits stop operating properly. This results, on the one hand, in stopping the data transmission from the transponder to the terminal and, on the other hand, in reducing the transponder remote-supply distance. 
     Another problem is that, to properly demodulate the received signal, a reader which is sufficiently sensitive to operate while voltage VD is close to VDMIN typically must be provided. This results in a risk of demodulator saturation when, conversely, voltage VD is higher, that is, when the transponder is very close to the reader. 
     SUMMARY 
     An embodiment of the present disclosure provides a novel solution which overcomes at least some disadvantages of conventional solutions, especially when a transponder is in remote or dose coupling relation with a read/write terminal. 
     In remote coupling relation, this embodiment reduces the supply voltage decrease effect linked to the distance between the transponder and the read terminal. 
     In close coupling relation, this embodiment reduces the disturbances of demodulation circuits of the terminal due to the distance between the transponder and the read terminal. 
     This embodiment also provides a solution which requires few or no structural modification of the transponder. 
     This embodiment of the present disclosure also provides a method for controlling by pulse trains a switch for modulating the load of a transponder in an electromagnetic field from which it extracts its power supply, the duty ratio of the pulses being controlled according to the transponder supply voltage. 
     According to an embodiment of the present disclosure, the control is linear. 
     According to an embodiment of the present disclosure, the control is by degrees. 
     According to an embodiment of the present disclosure, said supply voltage is compared with at least one minimum voltage. 
     According to an embodiment of the present disclosure, said supply voltage is compared with at least one maximum voltage. 
     An embodiment of the present disclosure also provides a transponder comprising a resonant circuit from which it extracts its supply voltage from an electromagnetic circuit radiated by a terminal, a pulse train modulation circuit having at least one switch modulating the load that it forms on the electromagnetic field, and means for controlling the duty ratio of the pulses according to said supply voltage. 
     According to an embodiment of the present disclosure, the control means are linear. 
     According to an embodiment of the present disclosure, the control means are digital. 
     According to an embodiment of the present disclosure, the duty ratio is conditioned by a comparison of said supply voltage with a minimum voltage. 
     According to an embodiment of the present disclosure, the duty ratio is conditioned by a comparison of said supply voltage with a maximum voltage. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Features and advantages of the present disclosure will be discussed in detail in the following non-limiting description of specific embodiments in connection with the accompanying drawings. 
         FIGS. 1 ,  2 A, and  2 B, previously described, are intended to show the state of the art and the problem to solve; 
         FIG. 3  shows in the form of blocks a first embodiment of the switch control method according to the present disclosure; 
         FIGS. 4A to 4G  illustrate in the form of timing diagrams the operation of the transponder of  FIG. 3  according to an embodiment of the disclosure; and 
         FIG. 5  shows in the form of logic gates an embodiment of the control function of the switch control according to the present disclosure. 
     
    
    
     DETAILED DESCRIPTION 
     Same elements have been designated with same reference numerals in the different drawings. For clarity, only those elements which are necessary to the understanding of embodiments of the present disclosure have been shown in the drawings and will be described hereafter. In particular, the digital electronic circuits have not been detailed on the reader side. 
     A feature of an embodiment of the present disclosure is to control the duty ratio of the control pulses of a transponder modulation circuit with the power supply thereof. 
     The duty ratio is obtained from a comparison of the supply voltage with at least one threshold. 
       FIG. 3  schematically shows an embodiment of a circuit for controlling the control signal of a back-modulation switch implementing the pulse train control method according to the present disclosure. For simplification, the rest of the transponder circuits have not been shown, and may be conventional. 
     In this example, a pulse train RMOD provided by control circuit  20  ( FIG. 1 ) is applied on a first input  23  of a circuit  26  (REGUL) for controlling the conduction of switch S ( FIG. 1 ). Circuit  26 , which is in practice comprised in circuit  20 , provides a signal ARMOD. Generally, the duty ratio of signal RMOD is set to 0.5. 
     Supply voltage VD of circuit  20  is applied to first respective input terminals of a comparator  21  and of a comparator  22 . Pulse train RMOD is applied on synchronization inputs of D-type flip-flops  24  and  25 . The outputs of flip-fops  24  and  25  form inputs, respectively VCH and VCL, of circuit  26 . Outputs COMPL and COMPH of comparators  21  and  22  are applied on inverting reset inputs R, respectively of flip-fops  25  and  24 . Voltage VD is applied to each of load inputs D of flip-flops  24  and  25 . On each rising edge of signal RMOD, voltage VD is charged respectively on outputs VCH and VCL and on first input terminals of comparators  22  and  21 . 
     A first so-called “low” limit voltage reference VDL and a second so-called “high” limit reference voltage VDH are respectively applied to the second input terminals of comparators  21  and  22 . Low voltage limit VDL is selected to be greater than minimum operation voltage VDMIN of transponder  10 . High limit VDH corresponds to a threshold from which the back-modulation level disturbs (saturates) the demodulation circuits of terminal  1 , on reading of the data transmitted by transponder  10 . A low level of signal COMPL causes the reset of flip-flop  25 . Similarly, a low level of signal COMPH causes the reset of flip-flop  24 . 
     Circuit  26  outputs a signal (ARMOD) for controlling switch S, having a conduction duty ratio controlled according to voltage VD. Preferably, signal ARMOD is active in the high state and controls the gate of switch S (generally, a MOS-type transistor). 
     As soon as voltage VD reaches one of the two low or high limits VDL or VDH, signals VCL or VCH provided by flip-flops  25  or  24  to circuit  26  cause, respectively, a reduction or an increase in the duty ratio of signal RMOD. At the output of circuit  26 , the duty ratio of signal ARMOD is either reduced under, or increased above, its nominal operation value, that is, without crossing limits VDL or VDH. As a result, the turning-off of switch S is either advanced in case low limit VDL has been crossed, or delayed in case high limit VDH has been crossed. 
     The variation range of the duty ratio is limited by two high and low thresholds. The low threshold is selected according to the conduction threshold of switch S short of which reader  1  can no longer correctly demodulate the data transmitted by transponder  10 . Generally, this low threshold is selected from a duty ratio range from 0.1 to 0.4. Symmetrically, the high threshold is selected according to the high conduction threshold of switch S beyond which reader  1  can no longer correctly demodulate the data transmitted by transponder  10 . Typically, this high threshold is selected from a duty ratio range from 0.6 to 0.9. 
       FIGS. 4A to 4G  illustrate, in the form of timing diagrams, an example of the shape of characteristic signals at different points of the embodiment described in  FIG. 3 .  FIGS. 4A to 4E  respectively show the signals of modulation RMOD, of voltage VD, of low and high comparison VCL and VCH, and of control signal ARMOD of switch S. These timing diagrams show, in their left-hand portion, a control of the conduction duty ratio of switch S resulting from a crossing of low limit VDL, and right-hand portion, a control resulting from a crossing of high limit VDH. 
     For a nominal operation with no crossing of one of limits VDL or VDH, the duty ratio of signal RMOD has been set to 0.5. As an example, thresholds DMIN and DMAX limiting the duty ratio range by minimum and maximum pulse durations have been respectively set to 0.25 and 0.75. These thresholds have been illustrated by two additional timing diagrams ( FIGS. 4F and 4G ) which are here for reasons of explanation only. In practice, the application of these thresholds requires no generation of specific pulse patterns, although, as discussed below in conjunction with  FIG. 5 , such patterns may be generated. 
     In an operation with a crossing of low limit VDL, assume that at a time t 1  during a conduction phase of switch S, voltage VD reaches low limit VDL. At this time t 1 , output VCL of flip-flop  25  switches to a low state. Assuming that time t 1  is under the minimum duration of a pulse, circuit  26  maintains signal ARMOD high. At a time t 2 , corresponding to the end of the minimum duration of a pulse, signal ARMOD switches and controls the turning off of switch S. The turning-off of switch S at time t 2  reduces the transponder consumption, causing an increase in voltage VD until a time t 3 . Time t 3  corresponds to the next turning-on of switch S triggered by circuit  26  at the rising edge of input signal RMOD. It is assumed that at a time t 4 , voltage VD reaches low limit VDL again, causing the premature turning-off of switch S. Here, time t 4  is subsequent to the minimum duration of a pulse set by threshold DMIN, which explains the immediate switching of signal ARMOD. From a time t 5  corresponding to the next turning-on, switch S is turned back on. Here, threshold VDL is not crossed. The duty ratio of signal ARMOD is thus at its nominal operation value (0.5 in this example). 
     It can be seen that the duty ratio of signal ARMOD is linearly controlled by circuit  26  as soon as voltage VD is smaller than VDL, while remaining limited to minimum threshold DMIN of the conduction duty cycle. 
     In an operation with a crossing of high limit VDH, assume that at a time t 6  during an off phase of switch S, voltage VD reaches high limit VDH. At this time t 6 , output VCH of flip-flop  24  switches to a high state. Time t 7  corresponds to the next turning-on of switch S triggered by circuit  26  at the rising edge of input signal RMOD. The nominal duration of the pulse started at time t 7  of signal RMOD ends at a time t 8 . The turning-on of switch S at time t 7  increases the transponder consumption, causing a decrease in voltage VD until a time t 9  for which it is assumed that voltage VD is again greater than high limit VDH. Since time t 9  is prior to the maximum duration of a pulse set by threshold DMAX, the switching to the low state of signal ARMOD and the turning-off of switch S are immediate. From a time t 10  corresponding to the next turning-on, switch S is turned back on. The turning-on of switch S at time t 10  increases the transponder consumption, causing a decrease in voltage VD until a time t 11  for which it is assumed that voltage VD is greater than high limit VDH. Here, time t 11  corresponds to the maximum duration of a pulse set by threshold DMAX, which explains the immediate switching of signal ARMOD. At a time t 12  corresponding to the next rising edge of input signal RMOD, the turning-on of switch S is triggered by circuit  26 . It is assumed that at a time t 13 , voltage VD reaches and falls under high limit VDH, causing the switching to the low state of voltage VCH. Assuming that time t 13  is within the minimum duration of a pulse, circuit  26  maintains signal ARMOD high until the end of the nominal duration of pulse RMOD. The duty ratio of signal ARMOD thus is at its nominal operation value (0.5 in this example). 
     It can be seen that the duty ratio of signal ARMOD is linearly controlled by circuit  26  on voltage VD as soon as it is greater than VDH, while remaining limited to maximum threshold DMAX of the conduction duty ratio. 
       FIG. 5  shows, in the form of logic gates, an embodiment of control circuit  26  ( FIG. 3 ) of signal RMOD. A first AND-type gate  31  receives on its two inputs signals DMAX and VCH. A second gate  32 , also of type AND, receives on its two inputs signals RMOD and VCL. Signal DMIN and the output signals of gates  31  and  32  are applied on three inputs of an OR-type gate  33 . The output of gate  33  corresponds to control signal ARMOD of circuit  26  of  FIG. 3 . 
     A digital embodiment in which the control means are formed by duty ratio stages may also be provided. These stages are organized in the form of a table stored in a non-volatile memory (not shown) of transponder  10  of  FIG. 1 . At its input, the table is formed of stages limited by a low and high value of voltage VD. Each high terminal of a stage corresponds to the low terminal of the stage directly above in terms of voltage VD. The low and high terminals of the stages located at the ends of the excursion range of voltage VD respectively correspond to voltages VDL and VDH of the first embodiment described in  FIG. 3 . In practice, it is assumed that voltage VD is first measured in analog form, then converted in digital form by any known analog-to-digital conversion means. At its output, the table provides a constant duty ratio for all the voltages VD ranging between the low and high terminals of a same stage. The duty ratios of the extreme stages of the excursion range of voltage VD respectively correspond to minimum and maximum thresholds DMIN and DMAX of the first embodiment described in  FIG. 3 . The resolution of the control depends on the number of stages selected to cover the excursion range of voltage VD, and on the granularity with which the digital conversion of analog signal VD is performed. 
     An advantage of an embodiment of the present disclosure is that it reduces the power sampled by the back-modulation circuit of the transponder in remote coupling relation. This results in increasing the remote-supply distance. 
     Another advantage is an improvement in the demodulation by the terminal for reading the data transmitted by the transponder, both in remote coupling and in dose coupling. An embodiment of the present disclosure enables increasing the sensitivity of the reader in remote coupling while strongly reducing the saturation risk of the reader&#39;s demodulator in close coupling. 
     Although embodiments of the present disclosure has been described in relation with the supply voltage, embodiments may be provided to control the duty ratio according to any other information or signal linked to this supply voltage. 
     Of course, the present disclosure is likely to have various alterations, modifications, and improvements which will readily occur to those skilled in the art. In particular, the sizing of the low and high limits of the transponder supply voltage, as well as of the minimum and maximum duty ratio thresholds, depends on the application and, in particular, on the frequency of the different carriers and on the system range. 
     Such alterations, modifications, and improvements are intended to be part of this disclosure, and are intended to be within the spirit and the scope of the present disclosure. Accordingly, the foregoing description is by way of example only and is not intended to be limiting.