Abstract:
The present invention provides an electric potential sensor for the measurement of potentials non-invasively. The sensor comprises at least one detection electrode arranged for capacitive coupling with a sample under test and for generating a measurement signal, and a sensor amplifier adapted to receive the measurement signal as input and to supply an amplified detection signal as output. Input impedance enhancing means are included for providing a high input impedance to the sensor amplifier for increasing the sensitivity of the electrode to reduced electric potentials, and a discrete pre-amplifier stage is arranged to co-operate with the sensor amplifier to reduce the input capacitance of the amplifier.

Description:
TECHNICAL FIELD 
     The present invention concerns electric potential sensors for use for the measurement of potentials non-invasively in a wide variety of applications, for example in the fields of medical diagnostics and biometric sensing. 
     BACKGROUND OF THE INVENTION 
     In order to create a sensitive electrodynamic measuring device, it is customary to provide a high input impedance and thereby reduce the power of the input signal required to operate the device. However, electronic circuits with a very high input impedance tend to be unstable, and so practical devices are usually a compromise between achieving the necessary degree of sensitivity, providing the desired input impedance and ensuring an acceptable degree of stability. 
     In International Patent Application No. WO 03/048789, an electrodynamic sensor is disclosed in which different circuit techniques are combined to achieve several orders of magnitude improvement in sensitivity, by comparison with previously known electrodynamic sensors, whilst still maintaining sufficient stability to permit a relatively unskilled operator to make measurements in everyday conditions. According to this earlier application, an electrodynamic sensor is provided, which comprises a high input impedance electrometer adapted to measure small electrical potentials originating from an object under test by means of at least one input probe, which has no direct electrical contact with the object. The circuit arrangement of the electrometer of this invention comprises an amplifier which includes a combination of ancillary circuits arranged cumulatively to increase the sensitivity of said electrometer to said small electrical potentials whilst not perturbing the electrical field associated therewith, the ancillary circuits serving to provide at least two of guarding, bootstrapping, neutralisation, supply rail drift correction, supply modulation and offset correction for said sensor. 
     Whilst these features assist in providing a sensor with high input impedance and a relatively stable operation, nevertheless, in situations where there may be weak capacitive coupling to, or a signal of small amplitude generated by, a source or sample under test, noise problems may still remain and may inhibit or prevent accurate signal measurement. This is particularly the case in certain medical and microscopic applications in which there is only a weak capacitive coupling and yet highly accurate signal measurement is essential, for example in a remote off-body mode of sensing in which the or each probe has no physical contact with the human body and typically the weak capacitive coupling would be &lt;1 pF. 
     More particularly, in applications where there is a weak coupling between a sample under test and the sensor electrode, the capacitive coupling to the sample may be comparable with or much smaller than the input capacitance of the sensor. In this case, the measurement signal received by the sensor is attenuated by the capacitive potential divider formed by the coupling capacitance and the input capacitance and may be difficult to capture. 
     There is thus a significant need for an electric potential sensor in which the possibility for accurate signal measurement is enhanced in cases of weak capacitive coupling to a sample under test. 
     Such a need is especially pronounced in cases where accuracy of signal measurement is crucial, for example in cases of biometric and medical measurement. 
     There is also a significant need for an electric potential sensor in which the signal to noise ratio is substantially improved. 
     BRIEF SUMMARY OF THE INVENTION 
     The present invention seeks to overcome the problems described above and to provide a novel electric potential sensor which is capable of highly accurate and non-invasive signal measurement. 
     The present invention, at least in the preferred embodiments described below, also seeks to provide an electric potential sensor in which the signal to noise ratio is significantly enhanced. 
     The present invention further seeks to provide various techniques and combinations of techniques for enhancing the signal to noise ratio in an electric potential sensor. 
     According to the invention, there is provided an electric potential sensor comprising:
         at least one detection electrode arranged for capacitive coupling with a sample under test and for generating a measurement signal;   a sensor amplifier adapted to receive the measurement signal as input and to supply an amplified detection signal as output;   input impedance enhancing means for providing a high input impedance to the sensor amplifier for increasing the sensitivity of the electrode to reduced electric potentials; and   a discrete pre-amplifier stage arranged to co-operate with the sensor amplifier to reduce the input capacitance of the amplifier.       

     According to the invention, the discrete pre-amplifier stage serves to increase the amplitude of the input measurement signal and thereby to increase the signal to noise ratio and enhance signal measurement. The discrete pre-amplifier stage may, for example, be provided by a high electron mobility transistor, or a FET arrangement. 
     The input impedance enhancing means may comprise at least one of a guard circuit, a bootstrapping circuit and a neutralisation circuit. The input impedance enhancing means may also further comprise one or more circuits for supply rail drift correction, supply modulation and offset correction for the sensor. 
     In a preferred embodiment described below, the detection electrode is juxtaposed with a conducting element connected to a zero reference potential in order to reduce effective source impedance, the conducting element being in the form of an annular ring surrounding the detection electrode. 
     In a further embodiment of the invention, there is provided in addition means for reducing the noise amplitude in order to increase the signal to noise ratio. For example, such means for reducing the noise amplitude may comprise at least one of a dc stability gain setting circuit, a noise matching circuit, and an enhanced bootstrap circuit. 
     The present invention thus aims to increase the signal to noise ratio either by increasing the amplitude of the signal or by decreasing the amplitude of the noise or both. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The present invention will now be described further, by way of example, with reference to the accompanying drawings, in which: 
         FIG. 1  is a circuit diagram of an electrodynamic sensor according to the prior art; 
         FIG. 2  is a block diagram of an electrodynamic sensor according to the present invention; 
         FIG. 3  is a circuit diagram of a first embodiment of a discrete pre-amplifier stage employed in the sensor of  FIG. 2 , with bootstrapping; 
         FIG. 4  is a circuit diagram of a modification of the  FIG. 3  circuit; 
         FIG. 5  is a circuit diagram of a further embodiment of discrete pre-amplifier stage provided by a FET, with bootstrapping and a DC level restorer circuit; 
         FIG. 6  is a circuit diagram of a modification of the  FIG. 5  circuit having a cascode circuit for bootstrapping the source of the FET; 
         FIG. 7  is a circuit diagram of a further modification of the  FIG. 5  circuit having a drain bootstrapping circuit for bootstrapping the drain of the FET; 
         FIG. 8  is a circuit diagram of a DC stability gain setting circuit shown in  FIG. 2 ; 
         FIG. 9  is a circuit diagram of a noise matching circuit shown in  FIG. 2 ; 
         FIG. 10  is a circuit diagram of a modification of the noise matching circuit of  FIG. 9 ; 
         FIG. 11  is a circuit diagram of an enhanced bootstrapping circuit shown in  FIG. 2 ; and 
         FIG. 12  is a circuit diagram of a modification of the enhanced bootstrapping circuit of  FIG. 11 . 
     
    
    
     DETAILED DESCRIPTION OF INVENTION 
     Referring to  FIG. 1 , an electrodynamic sensor as disclosed in International Patent Application No. WO 03/048789 will first be described. 
     As shown in  FIG. 1 , an eletrodynamic sensor  10  according International Patent Application number WO 03/048789 comprises a detection electrode  12  connected to the non-inverting input of a sensor amplifier  14 . In use, the detection electrode  12  supplies a measurement signal to the sensor amplifier  14 , whose output supplies an amplified detection signal as output. 
     The detection electrode  12  includes an electrode disc  16  mounted on a conductive stem  18 , the electrode disc  16  comprising a surface oxide layer  20  on a substrate  22 . The sensor amplifier  14  has a fixed input resistance  24 , connected between the electrode  12  and the non-inverting input of the amplifier  14 , to provide a steady input bias current to the amplifier  14 . In practice, the input resistor  24  will generally have a high resistance of the order of 100 GΩs or greater. The sensor amplifier  14  also has a guard  26  physically surrounding the input circuitry including the electrode  12  and the resistor  24  and providing a shield driven by the output of the amplifier  14 . Stray capacitance is thus alleviated by means of this positive feedback technique by maintaining the same potential on the guard or shield  26  as on the input detection electrode  12 . 
     In addition to the guard  26 , further circuit components may be provided for bootstrapping and neutralisation of the sensor as described in International Patent Application number WO 03/048789. 
     The earlier sensor shown in  FIG. 1  may be employed as a sensor probe for electrodynamic body sensing to obtain biometric measurements either in a contact mode, in which case the oxide layer  20  forms a capacitor providing relatively strong electrical coupling to the skin of a person under observation, or in an electrically isolated sensing mode, in which case the oxide layer  20  may be omitted and capacitive couple providing a relatively weak electrical coupling may be achieved through clothing or other intervening layers. 
     A sensor  28  according to the present invention will now be described with reference to  FIG. 2 , such sensor effectively comprising the sensor  10  of  FIG. 1 , with its detection electrode  12  and sensor amplifier  14 , but with the inclusion of further and different components to increase the accuracy of signal measurement, particularly in cases where a weak capacitive coupling to the subject under test is present. 
     One such additional component comprises an annular conducting element  12   a  surrounding the electrode  12  and connected to a reference voltage potential V r , such as earth or a zero potential point on the sensor amplifier  14 . The effect of the annular element  12   a  is to reduce the source impedance, ie the coupling impedance between the sample under test and the input of the sensor amplifier  14  as provided by a combination of coupling resistance R c  and coupling capacitance C c , by reducing the effective distance from the electrode  12  to the earthing point for the sensor amplifier  14 . The element  12   a  does not need to be annular but may have other configurations. 
     Further additional components are included in the circuitry of the sensor itself. More particularly, the sensor  28  of the present invention employs a discrete pre-amplifier stage  30 , having an intrinsically lower device input capacitance than is available in commercial operational amplifiers, in conjunction with the features of the sensor  10  of  FIG. 1 . Such a discrete pre-amplifier stage  30  is shown diagrammatically in  FIG. 2 , and embodiments of this discrete device are described with reference to  FIGS. 3 to 6 . The invention may also employ various bootstrapping techniques in conjunction with the pre-amplifier stage  30  in order to enhance the operation of the discrete device, as shown for example respectively in  FIGS. 2 to 6 . In addition, or instead, the invention may employ techniques for reducing the amplitude of the noise, as shown for example in  FIGS. 2 and 7  to  9 . 
       FIG. 2  is a block diagram of the sensor  28  according to the invention illustrating how these different techniques may be applied to the sensor to enhance significantly the signal to noise ratio. One embodiment of the discrete pre-amplifier stage  30 , which is in practice inserted between the detection electrode  12  and the sensor amplifier  14 , is shown in detail in, and further described in relation to,  FIGS. 3 and 4 . This embodiment includes a bootstrapping circuit  32  shown separately in  FIG. 2 . Another embodiment of discrete pre-amplifier stage  30 , also employing the bootstrapping circuit  32  as well as a DC level restorer circuit  34 , is illustrated in  FIG. 5 . 
     The discrete pre-amplifier stage  30  of  FIG. 5  may be further enhanced by means of additional bootstrapping, for example provided by a cascode circuit connection  36  as shown in  FIG. 6  to bootstrap the source of a FET employed as the pre-amplifier stage  30 , and/or provided by a drain bootstrap circuit  38  as shown in  FIG. 7  to bootstrap the drain of the FET. 
     Further techniques for noise reduction, illustrated in  FIGS. 8 to 12 , may also be applied to the sensor  28 . Such techniques may include the provision of a dc stability gain setting circuit  40  shown in  FIG. 8  for overcoming the problem of, low frequency instability, and/or of a noise matching circuit  42  shown in  FIGS. 9 and 10  for addressing the problem of low frequency noise, and/or of an enhanced bootstrap circuit  44  shown in  FIGS. 11 and 12  for addressing the problem of drift reduction. These additional circuits are all applicable generally to operational amplifier based sensors, as well as particularly to the versions of the sensor  28  including a discrete pre-amplifier stage  30  as described in relation to  FIG. 2 . In the case of weak coupling between the detection electrode  12  and a sample under test, however, the maximum signal to noise ratio will be obtained by utilising both a discrete pre-amplifier stage  30 , as described below with reference to  FIGS. 2 to 7 , and at least one of the techniques as described below in relation to  FIGS. 8 to 12 . 
     Discrete Pre-Amplifier Stage 
     For situations where the coupling capacitance C c  between a sample under test and the sensor  28  is much less than the input capacitance C in  of the sensor  28 , the available measurement signal is attenuated by a capacitive potential divider made up of the capacitances C c  and C in . This is the case in practice for many remote monitoring applications and for microscopic probes, particularly for example in the field of biometric sensing. In this situation, the best way of increasing the signal to noise ratio would be to reduce the input capacitance C in  to be less than or comparable with the coupling capacitance C c . However, commercially available operational amplifiers typically have input capacitances C in  ranging from 1-10 pF, and these cannot be reduced further. The present invention is based on the realisation that a discrete pre-amplifier stage  30  having an input capacitance as low as 0.1 pF may be employed in conjunction with the detection electrode  12  and sensor amplifier  14  effectively to achieve a lower input capacitance. The use of such a device as a front end pre-amplifier will increase the available signal by a large factor (×10-×100). 
     In one embodiment of the sensor  28  as shown in  FIG. 3 , the pre-amplifier stage  30  is achieved using a high electron mobility transistor (HEMT) device  50  situated between the detection electrode  12  of the known sensor, represented in  FIG. 3  by an input V in , and the operational amplifier  14  of the known sensor. The HEMT device  50  displays very low noise characteristics due to the extremely high mobility of the charge carriers in the semiconducting channel of the device. The HEMT device  50  is configured in this instance as a common source amplifier, with the property of inverting voltage gain. A resistor Rd limits the current flowing through the channel of the HEMT device  50 , and the DC operating point is set by the voltage applied to a gate resistor Rg connected to the gate of the HEMT device  50 . The output signal, taken from the drain of the HEMT device  50 , is amplified by an operational amplifier OPA 1 , constituting the sensor amplifier  14 , with the gain of the operational amplifier OPA 1  set by a feedback connection of two resistors R 1 , R 2  and a capacitor C 1 . 
     An attenuated version of the output from the operational amplifier OPA 1  is fed back and amplified by way of a positive feedback loop including the bootstrap circuit  32  (see  FIG. 2 ) comprising a further operational amplifier OPA 2  arranged to provide a bootstrap signal via a capacitor C 2  for the gate resistor Rg, thereby increasing the input impedance of the sensor  28 . The gain of the operational amplifier OPA 2  is set by two resistors R 3  and R 4 . In addition, a resistor R 5  provides a DC path for the input bias current required by the HEMT device  50 . 
     Further enhancement of the signal to noise ratio may also be achieved by physically separating the first stage transistor  50 , providing the pre-amplifier stage  30 , from the following electronics and operating at a reduced temperature, for example as shown in  FIG. 4 . The circuit of  FIG. 4  is similar to that of  FIG. 3 , with the exception that the portion of the circuit to the left of the dashed line is maintained at cryogenic temperatures for reduced temperature operation and the portion to the right is at room temperature. 
     It is to be noted that the HEMT device  50  may take the form either of a pre-amplifier in front of the sensor amplifier  14 , as shown in  FIGS. 3 and 4 , or else may be incorporated within the feedback loop of the following amplifier OPA 1 . Further, the HEMT device  50  may comprise two or more devices if used differentially. 
     In another embodiment shown in  FIG. 5 , a silicon dual gate MOSFET  60  (or two FETs so connected) is employed as the pre-amplifier stage  30 . The MOSFET  60  is biased by means of an appropriate drain resistor R d  to give an inverting voltage gain. The input signal from the sensor electrode  12  is coupled to gate G 2  of the MOSFET  60 , with gate G 1  of the MOSFET  60  being held at an appropriate bias voltage by means of a further resistor R G1 . An input bias current in this example is provided by a high value resistor R b , typically having a resistance in the range 10-100 GΩ, to which is connected the bootstrap circuit  32 , here comprising a parallel connection of a capacitor C 4  and resistor R 10  providing the necessary coupling and DC bias to the resistor R b . 
     The output of the MOSFET  60  in this embodiment, taken from the drain D, contains both the amplified input signal and an unwanted DC offset. This DC offset may be removed by means of the DC level restoring circuit  34  in conjunction with the following operational amplifier circuit OPA 3 , which is configured as a differential amplifier and which represents the sensor amplifier  14  of the sensor  28 . For this purpose, the gain of the operational amplifier OPA 3  is set by resistors R 6  and R 8  for its inverting input and by resistors R 7  and R 9  for its non-inverting input. In addition, a capacitor C 3  is connected across the resistor R 9 , so as to act as a low pass filter which rejects the AC component of the signal coupled to it, thereby leaving the DC offset. Hence, the difference signal, which is amplified by the operational amplifier OPA 3 , consists only of the wanted signal. This technique has the advantage that it responds to any DC drift present in the output of the MOSFET  60  and removes this from the signal below a corner frequency set by the time constant of the filter components. 
     The output from the operational amplifier OPA 3  is suitable to provide a positive feedback signal for the guard circuit as shown in  FIG. 1  and the bootstrap circuit  32  as already described, as well as a neutralisation circuit as described in International Patent Application No. WO 03/048789. An input capacitance&lt;1 pF for the sensor  28  with the configuration of  FIG. 5  has been measured in experimental trials using this embodiment. 
     It should be noted that the DC input bias current described above as being provided by the resistor R b  may in practice be provided by one or a combination of three means: First, by leakage through the bootstrap capacitor C 4  (usually the effective resistance of the capacitor is much lower than the resistance of the bias resistor); second, by the addition of the resistor R 10  in parallel with the bootstrap capacitor C 4 ; and third, by including a resistor to ground from the junction of the bias resistor R b  and the bootstrap capacitor C 4 . 
     The embodiments shown in  FIGS. 6 and 7  are variants of the circuit shown in and described with reference to  FIG. 5 , and these will now be described. Like parts are designated by the same reference signs, and will not be described further in detail. 
     In the version of the  FIG. 5  embodiment shown in  FIG. 6 , the silicon dual gate MOSFET  60  is connected in a cascode configuration where the device is internally bootstrapped to the source S so that internal bootstrapping is provided within the pre-amplifier stage. Such cascode connection  36  (see  FIG. 2 ) has the effect of greatly reducing the input capacitance C in  both of the MOSFET  60  and, since the MOSFET  60  is the input stage of the sensor  28 , of the overall sensor. For this circuit, the voltage gain of the first discrete pre-amplifier stage provided by the MOSFET  60  is unity and non-inverting. The output of the MOSFET  60  is again fed through the DC level restorer circuit  34  including the operational amplifier OPA 3  (amplifier  14 ) and is then coupled to an inverting amplifier OPA 4  to provide the correct phase of feedback signal for the bootstrap circuit  32 . The gain of the operational amplifier OPA 4  is set by two resistors R 11  and R 12 . A fraction of the output from the operational amplifier OPA 4  is used for the bootstrap circuit  32  as before. An input capacitance&lt;0.2 pF has been measured in experimental trials using this configuration for the sensor  28 . 
     A further enhancement of the  FIG. 6  embodiment with the cascode circuit connection  36  is possible as shown in the embodiment of  FIG. 7 . According to this embodiment, an additional bootstrap  38  to the drain D of the MOSFET  60  as well as the bootstrap to the source S enables the intrinsic input capacitance to be further reduced. This additional bootstrap  38  is achieved in this instance using a bootstrap capacitor C 5  connected between the MOSFET end of the parallel connection of the capacitor C 4  and resistor R 10  and the drain D of the MOSFET  60 . It is alternatively possible to employ an independently derived bootstrap signal obtained e.g. from the other end of the parallel connection of C 4 /R 10 . 
     By way of example, the input capacitance may be reduced to &lt;0.1 pF using the circuit of  FIG. 7 . This implies that so long as there is a coupling capacitance of ˜0.1 pF or greater, an optimum signal to noise ratio would be obtained. However, for this configuration of circuit, it is anticipated that the signal would remain measurable, with a 10:1 signal to noise ratio, for coupling capacitances down to ˜10 −15  F, assuming a 1 volt signal at the source. 
     The circuits of  FIGS. 3 to 7  significantly enhance the overall response of the electric potential sensor  28  to the sample under test in situations where weak coupling occurs to the sample. However, in certain circumstances, indicated below, problems still may arise at low frequencies of operation. The circuits shown in and described with reference to  FIGS. 8 to 12  address these problems. 
     DC Stability Gain Setting Circuit 
     The optimum noise performance of most amplifiers is achieved when the closed loop gain is considerably greater than unity, typically ×30-×100. Incorporating large voltage gain within the electric potential sensor  28  produces improvements in the noise performance, but may also introduce low frequency instability and increase the settling time of the sensor. One approach to alleviating this problem employs a low frequency negative feedback stabilisation loop as described in International Patent Application No. WO 03/048789. Another simple and effective technique is to introduce AC coupling into the gain setting network by employing a DC stability gain setting circuit  40  (see  FIG. 2 ) as shown in detail  FIG. 8 . Such DC stability gain setting circuit  40  may advantageously be employed in combination with one or more of the techniques described with reference to the embodiments of  FIGS. 3 to 7  but it may also offer benefits when employed alone in its own right. 
     More especially, the DC stability gain setting circuit  40  of  FIG. 8  comprises a series connection of a resistor R f  and a capacitor C f , between a negative feedback loop at the output of the sensor amplifier  14  of the sensor  28  and ground, for setting the time constant for lower frequencies of operation of the sensor  28 , where the time constant is given by:
 
 f   c =½π R   f   C   f  
 
     The effect of this is to reduce the gain of the sensor amplifier  14  to unity at DC whilst maintaining a high gain at the signal frequencies, hence stabilising the sensor and improving the settling time. Hence, it is possible to achieve low noise performance with high voltage gain and stability. 
     Noise Matching Circuit 
     The noise performance of a differential input amplifier, such as the sensor amplifier  14  of the sensor  28 , depends on many factors. Amongst the parameters to be considered are the level of the source impedance, ie the coupling impedance between the sample under test and the input of the sensor amplifier  14  as provided by a combination of coupling resistance R c  and coupling capacitance C c , compared to the input impedance, provided by a combination of input resistance R in  and input capacitance C in  for the amplifier  14 , and the extent to which the relative contributions of the voltage and current noise combine to create overall frequency dependent noise as observed at the output of the amplifier  14 . For a situation in which the coupling impedance between the sample and the input is very high (i.e. R c &gt;&gt;R in  and/or C c &lt;&lt;C in ), this factor may have a very large effect on the frequency dependent noise. 
     Close impedance matching between the inverting and non-inverting inputs of the sensor amplifier  14  serves not only to maximise the common mode rejection ratio, but also to minimise the noise. This may be achieved by the inclusion of a frequency dependent matching network, for example as shown in  FIG. 9 , providing the noise matching circuit  42  of  FIG. 2 . In this network, a parallel combination consisting of a resistor R m  and a capacitor C m , where R m =R c  and C m =C c , is added to the input of the sensor amplifier  14  to achieve this balance condition and hence a reduction in the frequency dependent noise observed at the output of the sensor amplifier  14 . 
     In a variation of the  FIG. 9  circuit, the parallel components R m , C m  could be replaced by a parallel combination of a FET and a varactor diode, with suitable biasing components, as shown in  FIG. 10 , to allow remotely tunable values for the resistance and capacitance for signal to noise optimisation. Bias voltages V g  and V v  control the resistance of the FET channel and the capacitance of the varactor diode respectively. 
     As in the case of the  FIG. 8  circuit, the noise matching circuit  42  of  FIG. 9  or  10  may advantageously be employed in combination with one or more of the techniques described with reference to the embodiments of  FIGS. 3 to 7  but it may also offer benefits when employed alone in its own right. 
     Bootstrap with Gain 
     The use of a positive feedback loop with a high pass characteristic to bootstrap the input bias network as described in International Patent Application No. WO 03/048789 significantly enhances the performance of the basic sensor  10  by increasing the input impedance. However, this technique may become difficult to implement at very low frequencies (say &lt;1 Hz) due to the long time constant required, as set by the values chosen for the resistor R and capacitor C of the bootstrap circuit. In other words, the signal to noise ratio is reduced at low frequency. One way of addressing this problem comprises the use of an enhanced bootstrap circuit  44  as shown in  FIGS. 2 and 11 , which utilises a higher gain output (e.g. ×10) available from the sensor amplifier  14 . For example, the provision of two gain setting resistors  9 R and R at the output of the amplifier  14 , signifying a 9:1 ratio for their resistance values, gives a gain of ×10. The bootstrap signal must then be precisely ×1 if maximum bootstrap and stable operation is to be achieved. In this enhanced bootstrap circuit  44 , the output signal from the amplifier  14  is fed back through the bootstrap capacitor C to a 1/10 resistive attenuator, comprising further resistors R and  9 R in a 9:1 ratio, shown on the left hand side of the capacitor C in  FIG. 11 , to provide the ×1 bootstrap signal. This results in a ×10 (for this example) increase in the time constant, therefore leading to smaller values for the capacitor C for a given lower operating frequency, or lower frequency operation. 
     A variation on the enhanced bootstrap circuit  44  of  FIG. 11  uses a high pass filter to set the lower operating frequency of the bootstrap circuit as shown in  FIG. 12 . Here, the time constant is set by two resistor-capacitor pairs (RC) connected in the feedback circuit from the output of the amplifier  14 , which RC pairs, together with a high impedance buffer amplifier OPA 6 , form a second order high pass filter. The gain is provided by two gain setting resistors R 1  and R 2 . 
     It will be appreciated that the variation of  FIG. 12  may employ either a passive high pass filter followed by a high impedance buffer amplifier or an active high pass filter, both of which enable low frequency operation to be achieved with convenient values of R and C. 
     Again, the enhanced bootstrap circuit  44  may advantageously be employed in combination with one or more of the techniques described with reference to the embodiments of  FIGS. 3 to 7  but it may also offer benefits when employed alone in its own right. 
     It should also be appreciated that the circuits described with reference to  FIGS. 8 to 12  may be employed individually or in combination with the techniques described with reference to the embodiments of  FIGS. 3 to 7 .