Abstract:
A X-Y addressable MOS imager sensor method and apparatus wherein a semiconductor based MOS sensor having an array of pixels forming the X-Y addressable MOS imager, the X-Y addressable MOS imager having a plurality of the pixels such that each pixel within the plurality of pixels has a photodetector with a reset mechanism that adjusts the photodetector potential to a predetermined potential level employs the measuring a plurality of reset levels with two different elapsed times between reset and measurement of the reset level, a comparison circuit operatively coupled to the means for measuring to determine a difference in reset levels, a predetermined set of transfer functions used to identify effective signal levels of the photodetectors, and determines from the difference which transfer function is applicable to that photodetector range of accumulated light. In response to the difference detected, transfer functions are applied to the charge read out from the photodetector. The transfer functions comprise a first transfer function that is pre-photodetector saturation function and a second transfer function that is a post-photodetector saturation transfer function. These transfer functions are applied to two reset levels of the photodetector within the same row readout period. The response to the difference can be the application of a series of digital adds and digital multiplies. The difference between the 2 reset levels is used to calculate the effective pixel signal level from the difference.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This is a divisional of U.S. Ser. No. 09/426,870, filed Oct. 26, 1999 now U.S. Pat. No. 6,486,504, which is related to U.S. application Ser. No. 09/426,833, filed Oct. 26, 1999, by Robert M. Guidash, and entitled, VARIABLE COLLECTION OF BLOOMING CHARGE TO EXTEND DYNAMIC RANGE. 
    
    
     FIELD OF THE INVENTION 
     The present invention pertains to semiconductor based image sensors and more particularly to Active Pixel image sensors (APS) or Passive Pixel image sensors (PPS) having increased dynamic range. 
     BACKGROUND OF THE INVENTION 
     APS are solid state imagers wherein each pixel contains both a photo-sensing means and at least one other active component, creating a charge that is converted to a signal (either a voltage or current signal). The signal represents the amount of light incident upon a pixel photosite. The dynamic range (DR) of an image sensing device is defined as the ratio of the effective maximum detectable signal level, typically referred to as the saturation signal, (V sat ), with respect to the rms. noise level of the sensor, (σ noise ) This is shown in Equation 1. 
     
       
         Dynamic Range= V   sat /σ noise   Equation 1 
       
     
     Image sensor devices such as charge coupled devices (CCD) that integrate charge created by incident photons have dynamic range limited by the amount of charge that can be collected and held in a given photosite, (V sat ). For example, for any given CCD, the amount of charge that can be collected and detected in a pixel is proportional to the pixel area. Thus for a commercial device used in a megapixel digital still camera (DSC), the number of electrons representing Vsat is on the order of 13,000 to 20,000 electrons. If the incident light is very bright and creates more electrons that can be held in the pixel or photodetector, these excess electrons are extracted by the anti-blooming means in the pixel and do not contribute to an increased saturation signal. Hence, the maximum detectable signal level is limited to the amount of charge that can be held in the photodetector or pixel. The DR is also limited by the sensor noise level, σ noise . Due to the limitations on Vsat, much work has been done in CCD&#39;s to decrease σ noise  to very low levels. Typically, commercial megapixel DSC devices have a DR of 1000:1 or less. 
     The same limitations on DR exist for APS devices. The V sat  is limited by the amount of charge that can be held and isolated in the photodetector. Excess charge is lost. This can become even more problematic with APS compared to CCD due to the active components within the pixel in the APS, limiting the area available for the photodetector, and due to the low voltage supply and clocks used in APS devices. In addition, since APS devices have been used to provide image sensor systems on a chip, the digital and analog circuits used on APS devices such as timing and control and analog to digital conversion, that are not present on CCD&#39;s, provide a much higher noise floor on APS devices compared to CCD. This is due to higher temporal noise as well as possibly quantization noise from the on-chip analog to digital converter. 
     Within the art of semiconductor based image sensors there are numerous disclosures that provide extended dynamic range for both APS and PPS devices. These include (1) measurement of number of clock periods to reach a threshold as taught by Konuma in U.S. Pat. No. 5,650,643; (2) capture of 2 or more correlated images with varying integration times as described by Orly Yadid-Pecht et al. in “Wide Intrascene Dynamic Range CMOS APS Using Dual Sampling” published in the 1997 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors; paper #R15; (3) logarithmic transfer function pixel architectures as described by Sypros Kavadias et al. in “On-chip Offset Calibrated Logarithmic Response Image Sensor”, 1999 IEEE Workshop on Charge-Coupled Devices, and Advanced Image Sensors, pp. 68-71, M. Loose et al., “Self-Calibrating Logarithmic CMOS Image Sensor with Single Chip Camera Functionality”, 1999 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors, pp. 191-194, N. Ricquier, et al., “Active Pixel CMOS Image Sensor with On-Chip Non-Uniformity Correction”, 1995 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors, . J. Huppertz et al., “Fast CMOS Imaging with High Dynamic Range”, 1997 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors, pp. 1-4; and (4) varying the level of the reset gate during integration as described in prior art references by S. Decker et al., “Comparison of CCD and CMOS Pixels for a Wide Dynamic Range Area Imager”, 1995 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors, and “A 256×256 CMOS Imaging Array with Wide Dynamic Range Pixels and Column-Parallel Digital Output”, IEEE Journal of Solid State Circuits, Vol 33, No. 12, December 1998. 
     U.S. Pat. No. 5,650,643 issued to Konuma (Konuma) teaches a device that can be used to increase the dynamic range of solid state image sensing devices. Konuma shows a means to increase the effective V sat  level by incorporating a comparator and a counter associated with a photodetector to measure the time required to reach an integrated signal threshold level, and provide this as the only sensor output. The counter is used in conjunction with the comparator to determine the number of counter clock periods that it took the photodetector to reach the signal level of that supplied to the comparator input. The device then provides only the number of counter clock periods as an output or signal value associated with the photodetector. 
     While the disclosure of Konuma does provide increased dynamic range through effectively increasing V sat , this approach has several problems. 
     First, if one were to have a counter and comparator in each pixel, the number of components in each pixel is very large leading to a pixel with a very small fill factor or a very large pixel. This approach is not practical given the present minimum feature sizes of state of the art semiconductor technology, and the need for small pixel, low cost image sensors. 
     Second, the output for each pixel is a counter value for the time required to reach a given threshold, and does not contain an analog output value for the actual amount of charge integrated in the photodetector. With this approach, although the effective V sat  level is increased, the effective DR will be limited by the time period or accuracy of the counter clock, and the size of the counter. For example, if the counter has 10 bits or 1024 counts, the dynamic range is extended to 10 bits provided that the 1024 master clock periods can fit into the desired exposure time. If the desired exposure time were 100 msec., then the counter clock period must be ≦97.6 usec. If one were try to extend the DR to 20 bits, a 20 bit counter would be required, and a counter clock frequency of &gt;10.5 MHz for an exposure time of 100 msec. In this example, the extension of the DR from 10 to bits results in a clock frequency requirement that is 1000 times faster. As the exposure time decreases a commensurately faster master clock is required. For example, if an exposure time of {fraction (1/60)} th  of a second were desired or required in the case of capturing an image in outdoors in bright sunlight, a master clock of 63 MHz would be required to quantize 20 bits. It is evident that very fast counter clocks are required to provide high dynamic range in typical exposure conditions. Also, as the number of bits in the counter gets larger, more area required to integrate this into the pixel, producing a larger and larger pixel. Typical counters require 4-8 transistors per bit. Thus a 20 bit counter would require 80-160 transistors, yielding pixel sizes of &gt;40 um in a 0.35 um CMOS process. Additionally this approach requires that all pixels within the image sensor reach the programmed threshold level in order to have an output value for each pixel. This would require very long exposure times to allow dark regions of the scene to reach the threshold level if the threshold level is near V sat . The exposure times could be decreased by programming the threshold level to a very low value, but this would reduce the accuracy of information in very bright regions of the scene since they will reach the threshold value in extremely short time periods. 
     Thirdly, with the approach of Konuma, at the brightest light levels the data is more quantized. This is shown is Equation 2 by looking at how the effective light measurement is calculated from the time to threshold. 
     If one knows the amount of time (t T ) required to reach a threshold (V T ) and assume that the source is constant over the time being measured, then one can calculate the amount of light at any arbitrary time, (t M ). The expression for the extended effective voltage (V ext ) is given by Equation 2 below.                V   ext     =           V   T     ·     t   M         t   T                    Equation                 2                                
     In a discrete system the time variable, t T , would be measured by a quantized unit as indicated in Equation 3.                t   T     =           t   M     ·   cv     MaxCv                  Equation                 3                                
     Where cv is the quantized integer code value and MaxCv is the code value that corresponds to the cv value at t M . Substituting values we arrive at Equation 4.                V   ext                =                      V   T     ·   MaxCv     cv                  Equation                 4                                
     Referring to FIG. 2, a code value (cv) of zero implies infinite light. The first measurable quantization, which is also the largest, is between cv=1 and cv=2. The quantization for an 8 bit linear system is 0.0039, which is less than the smallest quantization in a time to threshold method described by Konuma. 
     Fourthly, if one were to have a single counter and comparator used outside of the pixel array to keep track of the time to threshold, each pixel would then have to be measured at an extremely high rate in order to have a small enough sampling frequency per pixel to provide fine enough quantization for extension of the dynamic range. For example, assume that 10 bits of quantization over the desired exposure time is required, and that there are 1 million pixels in the image sensor. Given a desired exposure time of 100 msec., each pixel would have to be accessed and measured against the programmed threshold level every 97.65 usec. This means that 1 million pixels need to be sampled every 97.65 usec. This would require a pixel sampling rate of 1 pixel per 97.65 psec, or 10.24 GHz. A means for doing this is not disclosed by Konuma or elsewhere in the field of APS devices or other image sensor devices. 
     Finally, the output value provided is a time. In order to reconstruct the incident image from this output, (i.e. determine the signal level), one must extrapolate by multiplication from the time value. This can degrade the effective noise level of the sensor. The value t is used to measure the time for a voltage v(t) to reach to a threshold. The signal VPD(t) represents the accumulation of photons over time with some Gaussian additive noise with a standard deviation of σ v . One experienced in the art can show that the noise in the extended voltage domain (σ Ext ) is related to the additive noise as indicated by Equation 5.                σ   Ext     =         2   ·     σ   v     ·     t   M     ·     V   T   2           t   T     ·     (       V   T   2     -     σ   v   2       )                      Equation  5                                
     Given that t M  is always greater than t T  one can see that the value of σ Ext  is always greater than σ v . From the foregoing discussion it should be apparent that there remains a need within the prior art for a device that provides extended V sat  and dynamic range while retaining low noise, small pixel, simple and low frequency readout, and means to manage the quantization of extended voltage signals. 
     With the method of capturing two or more frames as disclosed by prior art references: Orly Yadid-Pecht et al. in “Wide Intrascene Dynamic Range CMOS APS Using Dual Sampling”; 1997 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors; paper #R15; O. Pecht et al., “CMOS APS with Autoscaling and Customized Wide Dynamic Range”, 1999 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors; and M. Schantz et al., “256×256 CMOS Imager with Linear Readout and 120 dB Dynamic Range”, 1999 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors, two or more frames of the same image or scene with different integration times are captured, multiple readouts are required and the integration times for each readout must be chosen in accordance with the scene illuminance in order to get an accurate interpolation. This adds complexity to the imaging auto-exposure system to facilitate rapid choice of integration times and has not been shown in the art. Also, additional frame memory is required to perform the multiple frame output comparison and calculation of effective signal level. Additionally, if there is any motion or change in scene illuminance between the 2 frames, this method of extending dynamic range will not work. 
     With the methods of extending dynamic range described by: Sypros Kavadias et al., “On-chip Offset Calibrated Logarithmic Response Image Sensor”, 1999 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors, pp. 68-71; M. Loose et al., “Self-Calibrating Logarithmic CMOS Image Sensor with Single Chip Camera Functionality”, 1999 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors, pp. 191-194; and N. Ricquier, et al., “Active Pixel CMOS Image Sensor with On-Chip Non-Uniformity Correction”, 1995 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors, a pixel with a logarithmic transfer function is provided. This approach of using a MOSFET sub-threshold exhibits behavior having very high fixed pattern noise. Approaches to correct this require extra system memory, individual threshold trimming of each pixel, or extra transistors per pixel. This increases chip size as well as system cost and complexity. 
     With the methods of extending dynamic described by J. Huppertz et al., “Fast CMOS Imaging with High Dynamic Range”, 1997 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors, pp. 1-4; and S. Decker et al., “Comparison of CCD and CMOS Pixels for a Wide Dynamic Range Area Imager”, 1995 IEEE Workshop on Charge-Coupled Devices and Advanced Image Sensors, the reset gate voltage level is changed from an on-state to an off-state with a specified time response during integration. With this method the dynamic range is extended by skimming off charge to the reset drain for bright pixels so that the pixel does not saturate. This method has the disadvantages of requiring complicated timing to operate in an electronic shutter mode, and difficulty in discerning whether or not a particular signal level is due to skimming of charge or is simply the total integrated signal level. 
     From the foregoing discussion it should be apparent that there remains a need within the prior art for a device that retains provides extended dynamic range while retaining low noise, small pixel, single frame readout, and flexible transfer function. 
     SUMMARY OF THE INVENTION 
     According to the present invention, there is provided a solution to problems of the prior art. In the present invention, blooming behavior of CMOS based image sensors is used to provide extended intrascene dynamic range. 
     A first embodiment of the present invention utilizes the charge that blooms from the photodetector during integration. In prior art devices the signal level of the photodetector is limited in bright regions by removing and discarding the blooming charge. In the present invention the charge that blooms from the photodetector is integrated for a period that is controlled independently from the photodetector integration time, and this collected blooming charge is added to the photodetector signal charge in the readout of the pixel. 
     A second embodiment of the present invention provides extended dynamic range by utilizing the behavior of diminishing signal with increasing light level associated with the readout of CMOS APS under high illumination levels. This is done by performing 2 samples of the reset level that have different elapsed times between reset and measurement of the reset level, thus providing 2 different effective reset level integration times. By comparing the 2 measured reset levels, one can determine the effective signal level of the photodetector beyond the physical saturation signal of the photodetector. The difference determined in the two reset measured levels is used to if the pixel is operating in the diminishing high light signal region, or in low light standard linear signal region. Depending on the result, a different transfer function is applied to voltage output from the from readout of the photodetector of that pixel to provide an effective signal level for that pixel. If the light that is incident on that pixel is bright, and a difference in the 2 measured reset levels exceeds a predetermined threshold, a transfer function associated with the diminishing signal region will be used to calculate the photodetector value. If a difference in the 2 measured reset levels does not exceed a predetermined threshold value, then the standard linear transfer function will be used to identify that photodetector value. Additionally, the value of the measured difference can be used independently to calculate or determine the effective signal level or incident illumination level for that pixel. 
     According to the present invention, an active pixel sensor device that significantly increases the dynamic range of the device and can be used in current system designs is provided by an X-Y addressable MOS imager sensor method and apparatus wherein a semiconductor based MOS sensor having an array of pixels forming the X-Y addressable MOS imager, the X-Y addressable MOS imager having a plurality of the pixels such that each pixel within the plurality of pixels has a photodetector with a reset mechanism that adjusts the photodetector potential to a predetermined potential level employs the measuring a plurality of reset levels with two different elapsed times between reset and measurement of the reset level, a comparison circuit operatively coupled to the means for measuring to determine a difference in reset levels, a predetermined set of transfer functions used to identify effective signal levels of the photodetectors, and determines from the difference which transfer function is applicable to that photodetector range of accumulated light. In response to the difference detected, transfer functions are applied to the charge read out from the photodetector. The transfer functions comprise a first transfer function that is pre-photodetector saturation function and a second transfer function that is a post-photodetector saturation transfer function. These transfer functions are applied to two reset levels of the photodetector within the same row readout period. The response to the difference can be the application of a series of digital adds and digital multiplies. The difference between the 2 reset levels is used to calculate the effective pixel signal level from the difference. 
     ADVANTAGEOUS EFFECT OF THE INVENTION 
     The invention has the following advantages. It provides for extending the dynamic range of a sensor that can easily be employed within current pixel and sensor architectures with little or no modification. Small pixel with high fill factor can detect signal level out to 40,000X V sat  single frame capture standard read-out. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a diagram of a first embodiment of the present invention that extends the dynamic range by collecting charge that bloomed from the photodetector; 
     FIG. 2 a  is diagram illustrating the operation of the pixel shown in FIG. 1 on a column basis where photodetector charge is added, and an associated timing diagram; 
     FIG. 2 b  is diagram illustrating the operation of the pixel shown in FIG. 1 on a column basis where photodetector voltage is added, and an associated timing diagram; 
     FIG. 3 a  illustrates a pair of transfer functions for the pixel in FIG. 1 operated by the timing diagrams shown in FIG. 2 having a short floating diffusion integration time leading to a small slope for linear region  2 ; 
     FIG. 3 b  illustrates a pair of transfer functions for the pixel in FIG. 1 operated by the timing diagrams shown in FIG. 2 having a relatively longer floating diffusion integration time leading to a larger slope for linear region  2 ; 
     FIG. 4 is a timing diagram used to illustrate the diminishing signal behavior shown in FIG. 5 a;    
     FIG. 5 a  is an illustration of a pixel cross section; 
     FIGS. 5 b  through  5   d  are electrostatic diagrams illustrating the effect of diminishing signal behavior for the pixel as illustrated; 
     FIG. 6 is a timing diagram used to illustrate the diminishing signal behavior shown in FIG. 7 a;    
     FIG. 7 a  is an illustration of a pixel cross section; 
     FIGS. 7 b  through  7   d  are electrostatic diagrams illustrating the effect of diminishing signal behavior for the pixel as illustrated; 
     FIG. 8 a  is a diagram illustrating the diminishing signal region; 
     FIG. 8 b  is an expanded diagram illustrating the diminishing signal region; and 
     FIG. 9 is a diagram illustrating the output o a sampled and held pixel. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The first embodiment of the present invention utilizes collection of charge that blooms from the photodetector during integration. This is accomplished using the pixel shown in FIG.  1 . This is a similar pixel design to that disclosed by Guidash et al in U.S. patent application Ser. No. 08/800,947. This pixel  10  comprises a photodetector  12  (preferably a pinned photodiode PD), a transfer gate TG  16 , a floating diffusion FD  18 , a light shield  8  over at least the floating diffusion  18 , an input signal transistor and a reset transistor comprised of source floating diffusion  18 , reset gate  17  and reset drain  19 . A microlens  6  is provided to focus light onto the photodetector  12 . Operation of the first embodiment of the present invention is shown in timing diagram of FIG. 2 for the pixel  10  of FIG. 1 resulting in the output signal transfer function shown in FIG.  3 . 
     Referring to FIG. 2 a , which is diagram illustrating the operation of the pixel shown in FIG. 1 on a column basis wherein charge from photodetector  12 , and floating diffusion  18 , is added by in the charge domain on the floating diffusion, with the combined charge from the photodetector and floating diffusion read out through amplifier  22 , and an associated timing diagram. The pixel  10  is reset or initialized by transferring charge from the photodetector  12  to the floating diffusion  18  by activation of the transfer gate  16  (shown as signal TG) and the subsequent resetting of the floating diffusion  18  by the activation of reset gate  17 , which resets the floating diffusion  18  to a predetermined potential. The photodetector integration time (t intpd ) commences when the transfer gate  16  is turned off after the initialization or reset sequence, at time τ 1 . The electrostatic potential of the transfer gate in its off state is set at a level deeper than 0 volts so that excess charge in the photodetector will bloom through the transfer gate region and into the floating diffusion  18 . As photodetector integration time (t intpd ) proceeds, the photodetector  12  will fill up with photoelectrons. If the incident light is bright, the number of photoelectrons generated in that pixel will exceed the capacity of the photodetector  12 . At this point the excess charge will spill into the floating diffusion  18 . If the reset gate  17  is on when charge begins to spill onto the floating diffusion  18 , this charge will be swept through the reset gate  17  region to reset drain  19  leaving the floating diffusion at the same predetermined potential, typically a threshold voltage below that of the reset drain  19  (VDD). If the reset gate  17  is off when charge begins to spill into the floating diffusion  18 , then this charge will be collected in the floating diffusion  18 . The amount of time that RG is turned off during the photodetector integration period is referred to as the floating diffusion integration time t intfd . 
     At the end of desired photodetector integration time t intpd , the signal charge in the photodetector  12  is transferred to the floating diffusion  18  by pulsing the transfer gate  16  on and off at time τ 2 . The signal level of the floating diffusion  18  is then sampled by SHS and held on capacitor C 1 . If no blooming charge has been collected on the floating diffusion  18  during the floating diffusion integration time t intfd ,(i.e. a dark region of the image), then the signal charge on capacitor C 1  from the floating diffusion  18  comprises the charge collected in the photodetector  12  during integration. If blooming charge has been collected on the floating diffusion  18 , then the signal charge on the capacitor C 1  from floating diffusion  18  comprises the charge collected in the photodetector  12  during photodetector integration time (t intpd ), plus the charge collected in the floating diffusion during the floating diffusion integration time. This floating diffusion integration time t intfd , is the time elapsed from when reset gate  17  is turned off to when the signal charge is transferred from the photodetector  12  at time τ 2 . The floating diffusion integration time t intfd  is controlled separately and independently from the photodetector integration time t intpd . 
     As a result of the operation described for FIG. 2 a  the sensor output response will be as shown in FIGS. 3 a  and  3   b . The output response comprises 2 regions. For low light levels the output response will follow linear region  1 . As the number of photoelectrons exceeds the capacity of the photodetector, this charge will be collected on the floating diffusion and added to the photodetector signal charge. In this case the pixel output response will follow region  2 . The preferred embodiment provides a linear response in region  2 , by the timing shown in FIG. 2 a . A non-linear response could be generated by having a time varying reset gate potential level as described in the prior art reference by S. Decker et al., “A 256×256 CMOS Imaging Array with Wide Dynamic Range Pixels and Column-Parallel Digital Output”, IEEE Journal of Solid State Circuits, Vol 33, No. 12, December 1998. In the preferred embodiment, the output response in region  2  is linear, and the slope is dependent on and directly proportional to the floating diffusion integration time t intfd . The two FIGS. ( 3   a  and  3   b ) illustrate the 2 different slopes for region  2 . The floating diffusion integration time in FIG. 3 a  is shorter than that for FIG. 3 b . Consequently, the slope of region  2  in FIG. 3 b . is greater than that for FIG. 3 a.    
     Since the floating diffusion integration time t intfd  is known, the effective signal level of the pixel can be determined by Equation 6. 
     
       
           V eff= V out for  V out&lt; V pdsat  Equation 6 
       
     
     
       
           V eff= V pdsat+( V out− V pdsat)( t   pdint   /t   fdint ) for  V out&gt; V pdsat 
       
     
     Since the ratio of the photodetector integration time t intpd  to the floating diffusion integration time t intfd  can be made large, Veff can be increased substantially over the signal limited by the photodetector capacity. Hence, the dynamic range is extended. Additionally, the maximum voltage level, Vmax, which is determined by the floating diffusion charge capacity can be larger than that of the photodetector capacity by controlling the capacitance of the floating diffusion. It is also possible to display the sensor output signal directly without determining the effective signal level from region  2 . This still provides extended instrascene dynamic range by mapping and directly displaying a larger incident illuminant range into the directly detectable signal voltage range. Also, the level of Vpdsat can be programmed by setting the off level of the transfer gate  16  to the desired level. 
     The timing shown in FIG. 2 a  shows a single readout of the total combined charge on the floating diffusion that comprises the photodetector signal and the collected blooming charge. The signal is read first followed by a reset and then a measurement of the reset level. This is an uncorrelated double sample readout, which adds temporal reset noise to the signal. It is possible to accomplish readout of the photodetector signal level and floating diffusion blooming charge level by performing two readouts. This is shown in FIG. 2 b . In this manner the level of blooming charge collected on the floating diffusion  18  is sampled and held by pulsing SHS fd  which places the blooming charge on Capacitor C 4 , followed by a reset of the floating diffusion  18  by RG and a sample and hold of the reset level by SHR which places the reset level on Capacitor C 5 . Charge is then transferred from the photodetector  12  to the floating diffusion  18  and that signal level is then sampled and held by SHS pd  and placed on Capacitor C 6 . The present invention envisions a differential readout for the stored blooming charge on capacitor C 4  and the reset level on capacitor C 5  via differential amplifier  31 , and a second differential readout for the stored signal charge on capacitor C 6  and the reset level on capacitor C 5  via differential amplifier  32 , thus providing true correlated double sampling for the photodetector signal level. The final output signal can then be determined by several means. One is reading the signals from the two differential amplifiers  31  and  32  separately providing two signal values per pixel that can be added off-chip. A second embodiment is accomplished by providing the signals as inputs to a 3 rd  differential amplifier and reading the signal out as a single level per pixel. This readout method of combining the signal in the voltage domain also provides a larger Vmax than combining the signals in the charge domain. This is because the floating diffusion does not have to hold the photodetector signal and blooming signal simultaneously. Hence the Vmax is extended to be the full floating diffusion capacity plus the photodetector capacity. 
     Since this method utilizes differential readout of the pixel with respect to a reference reset level, the pixel offset noise is cancelled. Additionally the dynamic range is extended without any additional components in the pixel, so that it can be accomplished with small pixels that are practical for low cost consumer digital imaging applications. With this approach pixel read noise will be increased by KTC due to the reset level being uncorrelated to the signal level. This will be typically less than 30 electrons and is small compared to the gain in effective signal level. 
     The second embodiment of the present invention also utilizes blooming behavior to extend dynamic range. In CMOS APS and PPS devices it is pointed out that as the incident light level becomes extremely bright, the output signal determined from the difference between the signal level and a reset level is perceived to decrease with increasing light levels, eventually reaching a “black” level. This diminishing signal behavior is due to the effect of the reset level decreasing as a result of charge accumulating on the charge to voltage conversion node, typically referred to as a sense node, during the time between reset and when the reset level is sampled and held. This occurs either by blooming charge from a separate photodetector being collected during on the sense node in the case of a four transistor APS, or by accumulation of electrons in the photodetector that also functions as the sense node in the case of three transistor APS pixels, or in the case of PPS pixels. If uncorrected, this diminishing signal effect with increasing incident light levels may be undesirable. 
     Cross sectional diagrams of the relevant region of pixels envisioned to operate under the second embodiment of the present invention are shown schematically in FIGS. 5 a  and  7   a  with corresponding timing diagrams of FIGS. 4 and 6. The pixel  20  shown in FIG. 5 a  and the corresponding timing diagram of FIG. 4 are for the case of either a three transistor APS device where the photodetector serves as the charge to voltage conversion node, and also or for the case of a PPS device. The pixel  30  shown in FIG. 7 a  and the corresponding timing diagram of FIG. 6 are for the case of an APS where the charge to voltage conversion node (floating diffusion  38 ) is separate and isolated from the photodetector  32 . 
     Referring to FIGS. 4 and 5 a , which is the case of three transistor APS devices and PPS devices, the pixel  20  is initialized by resetting the photodetector  22  when RG  27  is pulsed on and off. Integration of signal in the photodetector now commences. As integration proceeds the electrostatic potential of the photodetector VPD decreases until the photodetector  22  has completely filled up with photoelectrons. At this point the VPD is approximately 0 volts. The signal level of the photodetector  22  is then sampled and held by strobing SHS. Next the photodetector is reset by strobing RG  27  in order to provided a reference level differential readout of the signal from photodetector  22 . The time elapsed between the reset and sample and hold of the reset level is typically a few μsec. If the incident light level during this elapsed time is low, the reset level will not change appreciably and will be extremely close to the nominal reset level RL 0  as shown in FIG. 5 b . If this incident light is very bright, the reset level can change during this elapsed time if the sensor is not shielded from the incident light. The example seen in FIG. 5 c  is for a very bright incident light level that causes the photodetector voltage VPD to go to reset level RLA. Since the readout of the output signal level is determined by the difference between the signal level and the reset level, the perceived output signal level will decrease because the full photodetector level is compared against RLA rather than RL 0 . If the incident light level is bright enough to completely fill the photodetector in the time between reset and sample and hold of the rest level, the photodetector will reach a reset level RLB as seen in FIG. 5 d . In this case the difference between the sampled and held signal level and the sampled and held reset level is zero, producing a black or dark image. This produces a diminishing signal region with increasing incident light intensity as shown in FIGS. 8 a  and  8   b.    
     Referring to FIGS. 6 and 7 a , which is for the case of four transistor APS devices, the same effect occurs. As the incident light level becomes bright enough the photodetector  32  fills up and excess electrons spill into the floating diffusion  38 , consequently decreasing the reset level and the differentially read out signal level in a manner similar to that described for FIGS. 4 and 5 a  through  5   d.    
     However, measurements of this diminishing signal level effect show that the transfer function is very linear and reaches a black level at incident light levels that are 20,000× to 50,000× the normal saturation exposure, Isat, as shown in FIG. 8 b . In FIG. 8 a , the x-axis scale extends from no illumination to 3× times the saturation exposure. With this x-axis scale it is possible to discern the linear region output response of the pixel prior to saturation. This is the region between no illumination and Isat. The x-axis scale is changed in FIG. 3 b  to extend from no illumination to 50,000 times Isat. With this scale it is easier to discern the output response of the sensor in the saturated region, but difficult to discern the region prior to saturation shown in FIG. 3 a . In the output response of the sensor in the saturated region, the output signal diminishes linearly, or by some other characterized and empirically known transfer function, until reaching ˜50,000 times Isat, where the output signal reaches 0 volts and appears as a black or dark signal. 
     The new concept presented with the present invention is to use this diminishing signal region to determine the “effective signal level” to extend dynamic range. The “effective reset integration time” for this behavior is simply elapsed time between the falling edge of the reset of the charge to voltage conversion node and the falling edge of the sample and hold of the reset level, as shown in FIG.  4  and FIG.  6 . The signal change in this small integration time can be ascertained by having two reset sequences that are sampled and held with different effective integration times. If there is a difference between the two reset levels that exceed a predetermined threshold, then it is known that pixel is in the diminishing signal region of the response curve. This could also be done by comparing a single reset level to a reference reset level that is generated from a dark reset. Once it is determined that the pixel is in the diminishing region of the response curve, the incident illumination level can be determined from the specific measured transfer function provided by the diminishing signal region of the sensor. Thus incident illumination levels from no illumination up to 50,000× the saturation exposure can be detected by the sensor, providing extended dynamic range. Additionally, the effective signal level could be determined from the measured difference in the two reset level by using equation (7)                  V   eff     =       Δ                     V   reset          (       t   exp       t     int                 r         )         +     V   sat              
              where                   Δ                   V   reset       =       V     reset                 dark       -     V     reset                   illum   .                     Equation  7                                
     where t exp  exposure time and t intr =effective reset integration time. 
     Application of the saturated region transfer function, or calculation of the effective signal level as described in equation 7, could be accomplished outside of the image sensor via system software or hardware, or on the sensor in the analog or digital domain. One embodiment of this concept is shown in FIG.  9 . Referring to FIG. 9, a comparator with a programmable threshold is used to determine if there is a difference between the two reset levels with different elapsed times between reset and sample and hold of the reset level. The pixel output signal level is read in a manner similar to that described in prior art; the sampled and held photodetector signal level is read out differentially with respect to reset 1 , and digitized to N bits, (8 shown in this example). The output of this comparator with reset 1  and reset 2  as inputs is then used as a transfer function bit. A “zero” transfer function bit indicates that the 8 bit photodetector signal level should be used as is. A “one” transfer function bit indicates that the 8 bit photodetector signal level should be calculated using the 8 bit output using the saturated region transfer function. The timing for this readout operation is also shown in FIG. 9 for the pixel shown in FIG. 5 a . After the photodetector integration time has elapsed the floating diffusion is reset and the reset level is sampled and held. This is reset 1  and has an elapsed reset integration time  1 . Next the charge in the PD is transferred to the floating diffusion and the signal level is sampled and held. Next, the FD is reset and the reset level is sampled and held provided reset 2 , with an elapsed reset integration time  2  that is different than that for reset 1 . 
     The operation occurs in a similar manner for APS pixels where the PD functions as the sense node or for PPS pixels, except that the readouts occur in the sequence of read signal level, reset, read reset 1 , wait for an desired elapsed time, read reset 2 . 
     Additionally, rather than using a comparator to determine if there is a predetermined difference between reset 1  and reset 2 , the actual value of the differences in reset levels could be read out in a similar manner to that used to determine the value of (signal-reset 1 ). This value of (reset 2 -reset 1 ) could then be used to determine the effective signal level using equation 7. A comparison could be done to provide a flag bit to determine which value, either signal-reset 1 , or reset 2 -reset 1  to read out through the 8 bit ADC. 
     Parts List 
       10  pixel 
       12  photodetector 
       14  substrate 
       16  transfer gate 
       17  reset gate 
       18  floating diffusion 
       19  reset drain 
       20  pixel 
       22  photodetector 
       24  substrate 
       27  reset gate 
       28  floating diffusion 
       29  reset drain 
       20  pixel 
       22  photodetector 
       24  substrate 
       26  transfer gate 
       27  reset gate 
       28  floating diffusion 
       29  reset drain