Abstract:
An apparatus and method are provided. Generally, an input signal is applied across a main path (through an input network) and across a cancellation path (through a cancellation circuit). The cancellation circuit subtracts a cancellation current from the main path as part of the control mechanism, where the magnitude of the cancellation current is based on a gain control signal (that has been linearized to follow a control voltage).

Description:
TECHNICAL FIELD 
     The invention relates generally to variable gain amplifier and, more particularly, to a CMOS continuously variable gain amplifier. 
     BACKGROUND 
     Typically, variable gain amplifiers fall into one of two categories: multiplier based or digitally controlled. The multiplier based amplifiers are usually based on Gilbert cells and can provide continuous variability in terms of gain. Digitally controlled amplifiers, on the other hand, provide discrete gains, which may not be suitable for many applications. With conventional continuously variable amplifiers, such as Gilbert cell type amplifiers, step transitions (such as “on” and “off”) can generate distortion, such as a “pop” or “click” with audio applications. Additionally, Gilbert cell multipliers would generally require high performance bipolar transistors, which are not generally available in CMOS processes. 
     Turning to  FIG. 1 , an example of conventional continuously variable amplifier  100  can be seen. Amplifier  106  generally comprises a transconductance circuit  104 , a feedback network (i.e., resistors R 5  ad R 6 ), and a shunt circuit  106 . The shunt circuit  106  generally comprises resistors R 1  through R 4 , and shunt switches Q 1  and Q 2  (which can be NMOS transistors as shown). In operation, the gain control signal GC is applied to the shunt switches Q 1  and Q 2 . As the gain control signal GC is ramped or increased from 0V, the shunt switches Q 1  and Q 2  begin shorting the differential input signal applied to the shunt circuit  106  by the input source  102  so as to mute or “shut off” the input signal. Because this shunt circuit  106  is in the main signal path (from the input source  102  to the transconductance circuit  104 ), the shunt circuit  106  can introduce distortion. Point in fact, as the gain control signal GC increases, the total harmonic distortion (THD) increases, meaning that the THD performance of amplifier  100  limits the gain range of amplifier  100 . Therefore, there is a need for an improved amplifier. 
     Some other conventional circuits are: U.S. Pat. Nos. 5,436,588; 5,537,081; 6,774,684; and 7,227,413. 
     SUMMARY 
     A preferred embodiment of the present invention, accordingly, provides an apparatus. The apparatus comprises a transconductance circuit having a positive input terminal and a negative input terminal; an input network that is coupled to the positive and negative input terminals of the transconductance circuit and that receives an input signal; a control circuit that generates a gain control signal; and a cancellation circuit that is coupled to the positive and negative input terminals of the transconductance circuit, that is coupled to the gain control circuit, and that receives the input signal, wherein the cancellation circuit subtracts a cancellation current from the positive and negative input terminals of the transconductance circuit, and wherein the magnitude of the cancellation current is based at least in part on the gain control signal. 
     In accordance with a preferred embodiment of the present invention, the cancellation circuit further comprises: a first impedance network that receives a positive portion of the input signal that is coupled to the negative input terminal of the transconductance circuit; a second impedance network that receives a negative portion of the input signal and that is coupled to the positive input terminal of the transconductance circuit; and a plurality of shunt switches that are coupled between the first impedance network and the second impedance network, wherein each shunt switch is coupled to the control circuit so as to receive the gain control signal. 
     In accordance with a preferred embodiment of the present invention, the control circuit receives a control voltage and linearizes the gain control signal such that the gain control signal follows the control voltage. 
     In accordance with a preferred embodiment of the present invention, the control circuit further comprises: an amplifier having a positive input terminal, a negative input terminal, and an output terminal, wherein the negative input terminal of the amplifier receives the control voltage; an NMOS transistor that is coupled to the output terminal of the amplifier at its gate and the positive input terminal of the amplifier at its drain. 
     In accordance with a preferred embodiment of the present invention, the control circuit further comprises: a current source that is coupled to the drain of the NMOS transistor; and a resistor that is coupled between the drain and source of the NMOS transistor. 
     In accordance with a preferred embodiment of the present invention, the resistor further comprises a first resistor, and wherein the input network further comprises: a second resistor that receives the positive portion of the input signal and that is coupled to the positive input terminal of the transconductance circuit; and a third resistor that receives the negative portion of the input signal and that is coupled to the negative input terminal of the transconductance circuit. 
     In accordance with a preferred embodiment of the present invention, the first impedance network further comprises a plurality of fourth resistors coupled in series with one another, and wherein the second impedance network further comprises a plurality of fifth resistor coupled in series with one another, and wherein NMOS transistor further comprises a first NMOS transistor, and wherein the plurality of shunt switches further comprises: a second NMOS transistor that is coupled to the first impedance network at its drain and the output terminal of the amplifier at its gate; and a third NMOS transistor that is coupled to the second impedance network at its drain, the output terminal of the amplifier at its gate, and the source of the second NMOS transistor at its source. 
     In accordance with a preferred embodiment of the present invention, the amplifier further comprises a first amplifier, and wherein the transconductance circuit further comprises a positive output terminal and a negative output terminal, and wherein the apparatus further comprises: a second amplifier that is coupled to the positive output terminal of the transconductance circuit; and a third amplifier that is coupled to the negative output terminal of the transconductance circuit. 
     In accordance with a preferred embodiment of the present invention, a method is provided. The method comprises increasing a control voltage; generating a gain control signal that follows the control voltage; and subtracting a cancellation current from an input of a transconductance circuit, wherein the magnitude of the cancellation current is proportional to the gain control signal. 
     In accordance with a preferred embodiment of the present invention, the method further comprises amplifying the input signal that is applied to the input of the transconductance circuit through an input network. 
     In accordance with a preferred embodiment of the present invention, the input of the transconductance circuit further comprises a positive input terminal and a negative input terminal, and wherein the step of subtracting further comprises: applying a positive portion of the input signal to the negative input terminal of the transconductance circuit through a first impedance network; applying a negative portion of the input signal to the positive input terminal of the transconductance circuit through a second impedance network; and controlling a plurality of shunt switches coupled between the first and second impedance networks with the gain control signal. 
     In accordance with a preferred embodiment of the present invention, the step of generating the gain control signal further comprises setting the impedance of a control switch with a feedback loop such that gain control signal follows the control signal. 
     In accordance with a preferred embodiment of the present invention, an apparatus is provided. The apparatus comprises a positive input terminal for the apparatus; a negative input terminal for the apparatus, wherein the positive and negative input terminals for the apparatus receive positive and negative portions of an input signal, respectively; a first resistor that is coupled to the positive input terminal for the apparatus; a second resistor that is coupled to the negative input terminal for the apparatus; a cancellation circuit having: a third resistor that is coupled to the positive input terminal for the apparatus; a fourth resistor that is coupled to the negative input terminal for the apparatus; a plurality of shunt switches that are coupled between the third resistor and the forth resistor; a fifth resistor that is coupled to the third resistor; and a sixth resistor that is coupled to the fourth resistor; a control circuit that generates a gain control signal that follows a control voltage and that is coupled to provide the gain control signal to each of the shunt switches; and a transconductance circuit having a positive input terminal and a negative input terminal, wherein the negative input terminal of the transconductance circuit is coupled to the second and sixth resistors, and wherein the positive input terminal of the transconductance circuit is coupled to the third and fifth resistors. 
     In accordance with a preferred embodiment of the present invention, the plurality of shunt switches further comprises: a first NMOS transistor that is coupled to the third resistor at its drain and the control circuit at its gate; and a second NMOS transistor is coupled to the fourth transistor at its drain, the control circuit at its gate, and the source of the first NMOS transistor at its source. 
     In accordance with a preferred embodiment of the present invention, the control circuit further comprises: an amplifier having a positive input terminal, a negative input terminal, and an output terminal, wherein the negative input terminal of the amplifier receives the control voltage, and wherein the output terminal of the amplifier is coupled to the gates of the first and second NMOS transistors; a third NMOS transistor that is coupled to the output terminal of the amplifier at its gate and the positive input terminal of the amplifier at its drain; a seventh resistor that is coupled between the drain and source of the third NMOS transistor; and a current source that is coupled to the drain of the third NMOS transistor. 
     In accordance with a preferred embodiment of the present invention, the transconductance circuit further comprises a positive output terminal and a negative output terminal, and wherein the apparatus further comprises: an eight resistor that is coupled between the positive input and output terminals of the transconductance circuit; and an ninth resistor that is coupled between the negative input and output terminals of the transconductance circuit. 
     In accordance with a preferred embodiment of the present invention, the first, second, third, fourth, fifth, sixth, seventh, eighth, and ninth resistors are each about 100 kΩ, 100 kΩ, 90 kΩ, 90 kΩ, 10 kΩ, 10 kΩ, 10 kΩ, 100 kΩ, and 100 kΩ respectively. 
     In accordance with a preferred embodiment of the present invention, the transconductance circuit further comprises a positive output terminal and a negative output terminal, and wherein the amplifier further comprises a first amplifier, and wherein the apparatus further comprises: a second amplifier having a positive input terminal, a negative input terminal, and a positive output terminal, and a negative output terminal, wherein the negative input terminal of the second amplifier is coupled to the positive output terminal of the transconductance circuit; a third amplifier having a positive input terminal, a negative input terminal, and a positive output terminal, and a negative output terminal, wherein the negative input terminal of the third amplifier is coupled to the negative output terminal of the transconductance circuit, and wherein the positive input terminals of the second and third amplifiers receive a reference voltage; an eight resistor that is coupled between the positive input and output terminals of the second amplifier; and an ninth resistor that is coupled between the negative input and output terminals of the third amplifier. 
     In accordance with a preferred embodiment of the present invention, the first, second, third, fourth, fifth, sixth, seventh, eighth, and ninth resistors are each about 100 kΩ, 100 kΩ, 90 kΩ, 90 kΩ, 10 kΩ, 10 kΩ, 10 kΩ, 2 MΩ, and 2 MΩ, respectively. 
     The foregoing has outlined rather broadly the features and technical advantages of the present invention in order that the detailed description of the invention that follows may be better understood. Additional features and advantages of the invention will be described hereinafter which form the subject of the claims of the invention. It should be appreciated by those skilled in the art that the conception and the specific embodiment disclosed may be readily utilized as a basis for modifying or designing other structures for carrying out the same purposes of the present invention. It should also be realized by those skilled in the art that such equivalent constructions do not depart from the spirit and scope of the invention as set forth in the appended claims. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       For a more complete understanding of the present invention, and the advantages thereof, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which: 
         FIG. 1  is an example of a conventional continuously variable amplifier; 
         FIGS. 2 and 3  are examples of continuously variable amplifier in accordance with a preferred embodiment of the present invention; and 
         FIG. 4  is an example of a transconductance circuit of  FIGS. 2 and 3 . 
     
    
    
     DETAILED DESCRIPTION 
     Refer now to the drawings wherein depicted elements are, for the sake of clarity, not necessarily shown to scale and wherein like or similar elements are designated by the same reference numeral through the several views. 
     Turning to  FIG. 2 , an example of a continuously variable amplifier  200  can be seen. Amplifier  200  is similar in structure to amplifier  100 , but the shunt circuit  106  has been replaced with cancellation circuit  208 , control circuit  202 , and an input network. The control circuit  202  generally comprises amplifier  204 , a control switch Q 3  (which can be an NMOS transistor), resistor R 7  (which can have a resistance of about 10 kΩ and which is coupled to receive the common mode voltage VCM), and current source  206  (which is generally coupled to supply rail VDD). The input network generally comprises resistors R 8  and R 9  (which can each have a resistance of about 100 kΩ), and the cancellation circuit  208  generally comprises resistors R 1 , R 2 , R 3 , and R 4  (which can have resistance of about 90 kΩ, 90 kΩ, 10 kΩ, and 10 kΩ, respectively), and shunt switches Q 1  and Q 2  (which can be NMOS transistors). Additionally, resistors R 5  and R 6  of the feedback network can each have a resistance of about 100 kΩ. 
     A reason for including the control circuit  202  is to linearize the gain control signal GC. As shown, a control voltage VCNTL is applied to the negative input terminal of amplifier  204 . The amplifier  204 , resistor R 7 , and current source  206 , collectively, operate as a feedback or control loop for switch Q 3 . This control loop sets the impedance for switch Q 3  so that the voltage applied to the gate of switch Q 3  (which is the gain control signal GC) follows the control voltage VCNTL, effectively linearizing the gain control signal GC. 
     While the structure of the cancellation circuit  208  may appear to be similar to the shunting circuit  106 , there are some important differences. First, for amplifier  200 , there are two separate signal paths from the input source  102  to the transconductance circuit  104 , namely a main signal path through the input network and a cancellation path through the cancellation circuit  208 . Second, the impedance network (i.e., resistors R 1 /R 2 ) that carries the positive portion of the input signal is coupled to the negative input terminal of the transconductance circuit  104 , while the impedance network (i.e., resistors R 2 /R 4 ) that carries the negative portion of the input signal is coupled to the positive input terminal of the transconductance circuit  104 . This arrangement allows the cancellation circuit  308  to remove or subtract a cancellation current from the input terminals of the transconductance circuit  104  (instead of directly clamping), and this arrangement allows the distortion (introduced by cancellation circuit  208 ) to increase inversely with the control voltage VCNTL or gain control signal GC. Thus, the THD decreases proportionally with an increasing gain control signal, allowing for a wider gain range than amplifier  100  (i.e., a nearly 20 dB increase). 
     Turning to  FIG. 3 , another example of a variable gain amplifier  300  can be seen. Amplifier  300  is similar to amplifier  200 , but amplifier  300  also includes current-to-voltage converter  302  and  304 . Each of these voltage-to-current converters  302  and  304  includes an amplifier  306  and  308 , respectively, and a resistor R 11  and R 10 , respectively, (which can each have a resistance of about 2 MΩ. These current-to-voltage converters  302  and  304  generate output voltages OUTP and OUTM from the currents generated by the transconductance circuit  210 . 
     Turning to  FIG. 4 , the transconductance circuit  210  can be seen in greater detail. One feature of this transconductance circuit  210  is that it operates using different voltage domains (i.e., AVDD and PVDD). The supply voltage AVDD is typically a low voltage supply (i.e., 3.6V), while supply voltage PVDD is a high voltage supply (i.e., 200V). Generally, this arrangement can be used to drive motors or piezeoelectric transducers (i.e., for haptics applications). The transconductance circuit  210  generally comprises a level shifter  402 , input stages  406  and  408 , and output stage  404 . The transconductance circuit  210  also includes current bias network (having transistors Q 53  to Q 61  that receive bias currents IBIAS 1  and IBIAS 2 ) and transistors Q 40  through Q 42  (that can be included for reliability). Additionally, the level shifter  402  (which includes transistors Q 4  through Q 9 ) is generally a digital signal level shifter that generates a logic high signal at pump voltage VPUMP (which can be about 7.5V and that can be created by capacitive boost from supply voltage AVDD). 
     When the transconductance circuit  210  is “shut down” or off, the shut down signal SD is asserted or logic high, while enable signal EN is logic low. In this state, transistors Q 7 , Q 22 , Q 31 , Q 65 , and Q 71  are deactivated, while transistors Q 6 , Q 62  to Q 64  and Q 72  to Q 75  are activated. With Q 6 , Q 62  to Q 64  and Q 72  to Q 75  being activated, the level shifter  402  and current bias network are “off,” and nodes N 1  through N 7  are pulled to ground, which shuts off or inactivates input stages  406  and  408 . 
     When operational, the enable signal EN is logic high (while the shut down signal SD is logic low) so as to activate Q 7 , Q 22 , Q 31 , Q 65 , and Q 71  and to deactivate transistors Q 6 , Q 62  to Q 64  and Q 72  to Q 75 . In this operational state, the pump voltage VPUMP can then be provided via level shifter  402  to activate transistors Q 18  to Q 21 , which also activates the high voltage current mirrors (i.e., high voltage, isolated PMOS transistors Q 10  through Q 17 ) of the output stage  404 . The pump voltage VPUMP can also be applied to the current mirrors formed by transistors Q 23  to Q 26  so as to operate as a current source for the differential input pair Q 45  and Q 46  of input stage  406 . Additionally, because transistors Q 22  and Q 31  are activated, diode-connected transistor Q 27  is bypassed so as to apply supply voltage AVDD as the bias voltage for transistors Q 28 , Q 29 , Q 30 , and Q 32 , while the supply voltage AVDD is applied to the gates of transistors Q 36  and Q 37 . 
     In operation, the outputs from input stage  406  (i.e., nodes N 1  and N 2 ) are used to drive input stage  408 . As shown, the differential input signals INP and INM are applied to the gates of transistors Q 45  and Q 46  (differential input pair), which operates (in conjunction with transistors Q 49 , Q 51 , Q 67 , and Q 69 ) as a folded cascode amplifier. Additionally, the input stage  406  includes a common mode feedback circuit (which generally comprises transistors Q 50  and Q 68 ). Transistors Q 50  receives a common mode voltage VCM at its gate and generates a steering current for transistor Q 68  (which is diode-connected and which forms a current minor with transistors Q 67  and Q 69 ). The arrangement of the common mode circuit (as shown) is simpler than other conventional common mode feedback arrangements, which can, at least in part, be attributed to the inputs and outputs of input stage  406  being shorted so as to allow for a “built in” averaging. Additionally, the outputs from input stage  406  (i.e., nodes N 1  and N 2 ) are capacitively coupled to ground (i.e., through capacitors C 1  and C 2 ) 
     The outputs from the input stage  406  can then be used by input stage  408  to generate differential output signals HVOUTP and HVOUTM in the output stage  404 . Input stage  408  is generally comprised of a two portions (positive and negative) that include darlington pairs  410 ,  412 ,  414 , and  416 . As shown, each portion includes four source-followers (i.e., transistors Q 38 /Q 39 , Q 43 , Q 44 , and Q 52  for pairs  410  and  412  and transistors Q 33 /Q 34 , Q 35 , Q 47 , and Q 49  for the pairs  414  and  416 ). These source-followers are arranged such that the voltage on nodes N 1  and N 4  should be approximately equal to the feedback voltages FBM and FBP, respectively. Additionally, darlington pairs  410  and  414  are arranged as modified folded darlington pairs, where the source-followers (which are formed by transistors Q 38 /Q 39  and Q 33 /Q 34  and coupled to the output stage  404 ) are split to form two legs that are folded around current summing nodes N 3  and N 6 . This arrangement (in conjunction with Zener diodes D 1  through D 6 ) allows the currents at nodes N 3  and N 6  to be summed in the correct phases to obtain an AB current (i.e., for signals HVOUTM and HVOUTP) at nodes N 8  and N 9 . Thus, transconductance circuit  210  converts the input voltage (from signals INP and INM) to an output current (in signals HVOUTM and HVOUTP). 
     Having thus described the present invention by reference to certain of its preferred embodiments, it is noted that the embodiments disclosed are illustrative rather than limiting in nature and that a wide range of variations, modifications, changes, and substitutions are contemplated in the foregoing disclosure and, in some instances, some features of the present invention may be employed without a corresponding use of the other features. Accordingly, it is appropriate that the appended claims be construed broadly and in a manner consistent with the scope of the invention.