Abstract:
A power converter having a two stage boost circuit and a small boost converter. The main power flow for the power converter is via the two stage booster circuit having a single switch. The voltage spike of the switch is clamped by a diode and a capacitor. The energy at the capacitor is transferred to the power converter&#39;s output terminals by the small boost converter. The two stage boost converter topology enables the use of much lower voltage and Rdson MOSFET switches so as to reduce cost, switch conduction loss and turn on loss.

Description:
FIELD OF INVENTION 
   The present invention relates in general to DC-DC converters, and more particularly, to a boost converter circuit topology for high power applications that utilizes a second smaller boost converter for reducing conduction loss and enabling the use of a substantially lower voltage rated MOSFET main boost switch. 
   BACKGROUND OF INVENTION 
   A boost converter is a power converter where a smaller input DC voltage is increased to a desired level. A prior art typical boost converter  10  is shown in  FIG. 1 . Boost converter  10  has input terminal  2 ,  4  for enabling an input voltage Vin to be coupled to converter  10 , and output terminals  6 ,  8  where the output DC voltage is provided. The boost converter  10  includes an inductor  18 , to which the input voltage Vin is coupled, that is in series with a boost diode  16  connected to an output capacitor  12  across which the load (not shown) is connected at terminals  6 , 8 . A transistor switch  14  is connected to a node  15  between the inductor  18  and boost diode  16  and a ground return line  20  to provide regulation of the output voltage. The switch  14  is typically a MOSFET having a control input, a drain and a source terminal. A control circuit  42  (details not shown) is coupled to the control input for providing a control signal for controlling the timing of the on and off transition of the switch  14 . The control circuit  42  typically has a pulse width modulation circuit (PWM). In operation, when the switch  14  is on, the inductor current increases, storing energy in its magnetic field. When the transistor switch  14  is off, energy is transferred via the diode  16  to the output capacitor  12  and the load. Drawbacks of such conventional boost converter circuits include the creation of switch voltage and current stresses resulting in low efficiency power conversion. 
   In a conventional boost converter, the voltage rating required for the MOSFET  14  is determined by the voltage appearing across the MOSFET when the control circuit  42  turns it off since the inductor  18  rises up to the output voltage plus any overshoot. A conventional boost converter with an output voltage of 250 VDC requires a MOSFET having a voltage rating (drain to source) of at least 400 volts. Conventional boost converter using in high step-up ratio applications, boosting 30V dc to 250V dc for example, therefore, require a high current and high voltage rated MOSFET. A drawback of the use of high current and high voltage rated MOSFETs is increased size and cost. 
   The voltage drop across a MOSFET  14  between the drain and source terminal is a function of the resistance (Rdson) provided that the load current is constant. Conduction losses for a MOSFET are equal to I 2 R losses, therefore, the total resistance between the source and drain terminals during the on state, Rdson, should be as low as possible. Consequently, a drawback of converters requiring higher Rdson rated MOSFETS is higher conduction losses. 
     FIG. 2  shows a prior art boost converter  20 . The boost converter  20  adds a snubber circuit  24  to the boost converter  10  of  FIG. 1 . The snubber circuit  24  is designed to absorb energy from the leakage inductance in the circuit and transfer this energy to the output. The snubber circuit  24  includes a capacitor  22  connected in series between inductor  18  and diode  36  and ground return line  20 . Snubber circuit  24  also includes a series combination of another inductor  28  and another diode  26  connected between the junction of the capacitor  22  and the diode  36 . 
   In operation, when the MOSFET  14  in boost converter  20  switches from ON to OFF, the voltage at node  25  is clamped by capacitor  22  and diode  36  to terminal  6  before diode  16  is turned ON. As a result, the voltage at node  25  will be slightly higher than the output voltage at terminal  6 . This causes diode  16  to turn ON and clamp the voltage of node  25  to Vout. When MOSFET  14  is ON, the charge (from the leakage energy) stored at capacitor  22  will flow through inductor  28  and diode  26 . At that moment, capacitor  22  and inductor  28  form a resonant network that reverses the polarity of capacitor  22 . More specifically, before MOSFET  14  is turned ON, the end of capacitor  22  at node  25  is positive, but becomes negative after MOSFET  14  turns ON and the resonant action completed. Thus, before MOSFET  14  turns OFF, node  25  is negative relative to the other end of capacitor  22 . As a result, the voltage spike at node  25  is clamped more effectively. 
   A drawback of converter  20  is that it is not suitable for high power or high boost ratio applications. More specifically, in high power applications or high boost ratio applications, the switching current at MOSFET  14  becomes high and the added capacitor  22  must be able to handle the high current demands, making it more difficult to find a suitable capacitor for a particular application. Moreover, in high boost ratio applications, a high current and high voltage rated MOSFET  14  is still required for converter  20  since the MOSFET  14  drain terminal is still clamped to the output of the boost converter. Although an improvement over converter  10  of  FIG. 1 , converter  20  has the similar drawback of having an unacceptably high voltage spike (&gt;Vout) at the drain of MOSFET  14  for high current applications so as to require higher voltage rated MOSFETS with higher Rdson and corresponding higher conduction losses. 
   A need therefore exists for a boost converter topology for high power applications to enable the use of lower voltage and lower Rdson rated MOSFETs and to provide increased efficiency by reducing conduction losses. 
   SUMMARY OF THE INVENTION 
   The aforementioned and related drawbacks associated with prior art boost converters are substantially reduced or eliminated by the improved boost converter topology of the present invention. 
   The present invention overcomes the drawbacks of known boost converters by providing a power converter having a two stage boost circuit and a second smaller boost converter. The main power flow is handled by the two stage boost circuit having a single main switch. The voltage spike, Vds, of the main switch is clamped by a first diode and a first capacitor. A second boost converter transfers the energy of the first capacitor to the output terminal, V out , and keeps the voltage constant across the first capacitor. The second boost converter is only required to handle the energy of the voltage spike at the first switch; consequently the second boost converter is much smaller than the main two stage boost circuit. The power converter of the present invention enables the use of lower voltage and lower Rdson rated MOSFETs for high power applications so as to provide increased efficiency by reducing conduction losses. 
   Broadly stated, the present invention provides a power converter having first and second input terminals where an input voltage is provided and two output terminals where the output DC voltage is provided comprising a two stage boost circuit comprising a first stage circuit comprising a first switch alternately switched on and off as a function of a first control signal for controlling current through a first winding of a magnetically coupled inductor, the first winding having one end connected to the first input terminal, a first diode, a first capacitor, the first switch coupled between the junction of the other end of the first winding and the first diode and the second input terminal, and a second stage circuit comprising a second winding of the magnetically coupled inductor connected to the other end of the first winding and connected in series with a second diode between the first winding and the first output terminal, and a second capacitor connected across the output terminals; and a second boost converter coupled between the first capacitor and the second capacitor comprising a second switch, an inductor, and a third diode; the second switch is connected between the junction of the inductor and the third diode and the second input terminal and is alternately switched on and off as a function of a second control signal. 
   Broadly stated, the present invention also provides in an alternate embodiment a power converter having first and second input terminals where an input voltage is provided and two output terminals where the output DC voltage is provided comprising: a two stage boost circuit comprising a first stage circuit comprising a first switch alternately switched on and off as a function of a first control signal for controlling current through a first winding of a magnetically coupled inductor the first winding having one end connected to the first input terminal, a first diode, and a first capacitor, and a primary winding of a current transformer connected in series between the first winding and the junction of the first switch and the first diode, the first switch coupled between the junction of the current transformer primary winding and the first diode and the second input terminal, and a second stage circuit comprising a second winding of the magnetically coupled inductor connected to the other end of the first winding and connected in series with a second diode between the first winding and the first output terminal, and a second capacitor connected across the output terminals; a second boost converter coupled between the first capacitor and the second capacitor comprising a second switch, an inductor, and a third diode; the second switch is connected between the junction of the inductor and third diode and the second input terminal and is alternately switched on and off as a function of a second control signal; and a current sensor for sensing the current through the first switch, the current sensor including a current transformer formed by the primary winding and a secondary winding, wherein the current transformer secondary winding is connected to a measurement resistor through a fourth diode. 
   An advantage of the present invention is that it enables the use of much lower voltage and Rdson MOSFET switches that are smaller and less costly, and reduces switch conduction and turn on losses. 
   Another advantage of the present invention is that it is suitable for use in high step up ratio applications. 
   These and other embodiments, features, aspects, and advantages of the invention will become better understood with regard to the following description, appended claims and accompanying drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The foregoing aspects and the attendant advantages of the present invention will become more readily appreciated by reference to the following detailed description, when taken in conjunction with the accompanying drawings, wherein: 
       FIG. 1  is a schematic diagram of a power converter having a conventional boost topology; 
       FIG. 2  is a diagram of a boost converter topology including a snubber circuit. 
       FIG. 3  is a schematic diagram of a preferred embodiment of the boost converter topology according to present invention; 
       FIG. 3A  is a schematic diagram of an alternate embodiment of the boost converter topology according to the present invention that include a current sense circuit; 
       FIG. 4  is an exemplary timing diagram that illustrates the current through the first and second windings and an exemplary control signal applied to the first switch of the converter  100  in  FIG. 3 ; 
       FIG. 5  is a timing diagram that illustrates voltage waveforms at various points for the boost converter in  FIG. 3  operating with an exemplary input voltage of 30VDC and an output voltage of 250VDC at 2400 W; and 
       FIG. 6  is a timing diagram that illustrates voltage waveforms at various points for the boost converter in  FIG. 3 . 
   

   Reference symbols or names are used in the Figures to indicate certain components, aspects or features shown therein, with reference symbols common to more than one Figure indicating like components, aspects or features shown therein. 
   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 3  is a schematic diagram of a preferred embodiment of the power converter  100  according to present invention. In a preferred embodiment, the power converter  100  has a boost converter topology including first  102  and second  104  input terminals where an input voltage is coupled and two output terminals  106 ,  108  where the output DC voltage is provided. Power converter  100  includes a two stage boost circuit  110  and a second boost converter  130 . The two stage boost circuit  10  has a first stage circuit  180  and a second stage circuit  170 . The two stage boost circuit  110  includes a magnetically coupled inductor  122  having a first winding  118  having one end connected to the first input terminal  102  and a second winding  120  connected to the other end of the first winding  118 . 
   The first winding  118  of magnetically coupled inductor  122  is part of the first stage circuit  180 . The first stage circuit  180  also includes a first switch  114  having a control input. The switch  114  is typically a MOSFET having a gate, a drain, and a source terminal. A control circuit  142  (details not shown) is coupled to the control input for providing a first control signal  146  for controlling the timing of the on and off transition of the first switch  114 . The control circuit  142  preferably includes a conventional PWM controller. Alternatively, the conventional PWM controller also provides power factor correction. The control circuit  142  preferably operates using conventional voltage mode control responsive to the output voltage to control the duty cycle of first switch  114 . The first switch  114  is alternately switched on and off as a function of the first control signal  146  for controlling current through the first winding  118  of the magnetically coupled inductor  122 . The first stage circuit  180  includes a first diode  116  having a cathode and an anode. The anode of the first diode  116  is connected to the junction of the first switch  114  and the other end of the first winding  118 . As shown in the embodiment in  FIG. 3 , the drain terminal of first switch  114  is connected to the junction of the anode of the first diode and the first winding  118  at a node  115 . The source terminal of the first switch  114  is connected to a ground return line  138  connected to the second input terminal  104 . The first stage circuit  180  includes a first capacitor  112  connected between the cathode of the first diode  116  and the second input terminal  104 . 
   The second stage circuit  170  of the power converter  100  includes the second winding  120  of the magnetically coupled inductor  122 . One end of the second winding  120  is connected to the end of the first winding  118  that is not connected to the input terminal  102 . The second stage circuit  170  includes a second diode  126  and a second capacitor  124 . The second diode  126  has an anode connected to the other end of the second winding  120  and a cathode connected to the first output terminal  106 . The second capacitor  124  is connected across the output terminals  106 ,  108  and has one end connected to the junction of the cathode of the second diode  126  and the first output terminal  106  and the other end connected to the second input terminal  104  and second output terminal  108 . 
   The number of turns in the first winding  118  and second winding  120  of the magnetically coupled inductor  122  need not be the same for practicing the present invention. The number of turns is a design choice as a function of the input voltage, output voltage, and the boost MOSFET used. In a preferred embodiment, the turn ratio of first winding  118  to second winding  120  is 12:16. The output voltage can be increased by increasing the number of turns in the second winding  120  while keeping the number of turns of the first winding  118  unchanged. The RMS current flowing through the first winding  118 , however, will also be increased. Consequently, the turn ratio between the first winding  118  and the second winding  120  is to be adjusted for the particular application so as to get the highest efficiency. 
   The second boost converter  130  is coupled between the first capacitor  112  and the second capacitor  124 . The second boost converter  130  includes a second switch  134 , an inductor  132 , and a third diode  136  having an anode and a cathode. The second switch  134  is typically a MOSFET having a control input, a drain and a source terminal. A second control circuit  144  is coupled to the control input for providing a second control signal  148  for controlling the timing of the on and off transition of the second switch  134 . The second switch  134  is alternately switched on and off as a function of the second control signal  148  for controlling current through the inductor  132 . The second control circuit  144  (details not shown) preferably includes a conventional PWM controller for simple voltage mode control. As shown in the embodiment in  FIG. 3 , the drain terminal of second switch  134  is connected to the junction of the anode of the third diode  136  and one end of inductor  132 . The source terminal of the second switch  134  is connected to the second input terminal  104 . The other end of inductor  132  is connected to the junction of the first capacitor  112  and the first diode  116 . Inductor  132  is connected in series with the third diode  136  between the junction of the first capacitor  112  and the first diode  116  and the first output terminal  106 . The source terminal of the second switch  134  is connected to the second input terminal  104 . The cathode of the third diode  136  is connected to the junction of the first output terminal  106  and the second capacitor  124 . 
   The operation of power converter  100  is explained in further detail with reference to  FIGS. 3–5 . In operation, when the first switch  114  is ON, current flows from the input terminal  102  through the first winding  118  and the first switch  114 . Energy is stored into the first winding  118 . Switching the first switch  114  OFF causes current to flow through the first winding  118 , the second winding  120 , and the second diode  126  to the output terminal  106 . As a result, when the first switch  114  is OFF, energy is released by the first winding  118  and the second winding  120 .  FIG. 4  is an exemplary timing diagram that illustrates the current through the first and second windings and an exemplary control signal applied to the first switch  114  of the converter  100  in  FIG. 3 . In  FIG. 4 , waveform A 1  is the first control signal  146  at the control input (gate) of switch  114 . Waveform A 2  is the current at the second winding  120 . Waveform A 3  is the current at the first winding  118 . The average current I avg  and the peak current I p  for waveform A 3  for the first winding  118  are also shown. For the waveforms in  FIGS. 4 , perfect coupling is assumed between the first and second windings of the magnetically coupled inductor  122 . 
   When the first switch  114  is OFF, a voltage spike, V ds , from the drain to source terminals of the first switch  114  is clamped by the first capacitor  112  through first diode  116 . The energy at the first capacitor  112  is transferred to the output terminals by the second boost converter  130 . For power converter  100 , the second boost converter  130  may be very small since it is required to handle the energy of the voltage spike, V ds , only. 
     FIG. 5  is a timing diagram that illustrates voltage waveforms at various points for the boost converter in  FIG. 3  operating with an exemplary input voltage of 30VDC and an output voltage of 250VDC at 2400 W. In  FIG. 5 , waveform B 1  is the voltage at node  115  at the junction of the first winding  118  and second winding  120  of magnetically coupled inductor  122 . As shown in  FIG. 5 , the maximum voltage at node  115  is 160VDC. Waveform B 2  is the voltage at the junction of the second diode  126  and the second winding  120 , that is, at the anode of the second diode  126 . As can be seen in  FIG. 5 , the voltage at the anode of the second diode  126  reaches the exemplary 250VDC output voltage. Waveform B 3  is the voltage of the first capacitor  112 . As can be seen in  FIG. 5 , the voltage of the first capacitor  112  averages 145 volts. The first control circuit  142  that provide the first control signal  146  for switch  114  and the second control circuit  144  that provides the second control signal  148  for second switch  134  are preferably separated. 
     FIG. 6  is a timing diagram that illustrates voltage waveforms at various points for the boost converter in  FIG. 3  including points for the small boost converter  130 . In  FIG. 6 , waveform C 1  is the gate drive control signal  146  for the first switch  114 . Waveform C 2  is the gate drive control signal  148  for the second switch  134 . The control signals for the first switch  114  and second switch  134  are preferably synchronized so the output ripple and noise will be regular and generally lower. Alternatively, the control signals are not synchronized. Waveform C 3  is the drain current through the second switch  134 . Waveform C 4  is the current through inductor (choke)  132 . 
     FIG. 3A  is an alternate embodiment of the power converter according to the present invention including a current sense circuit. Power converter  200  in  FIG. 3A  includes a two stage boost circuit  210  and a second boost converter  130 . The two stage boost circuit  210  has a first stage circuit  280  and a second stage circuit  170 . Power converter  200  includes a current sensor  160  having a current transformer  150  having a primary winding  152  and a secondary winding  154 . The current transformer primary winding  152  is part of a first stage circuit  280 . The anode of the first diode  116  is connected to the junction of the first switch  114  and the primary winding  152  of a current transformer  150 . As shown in the embodiment in  FIG. 3A , the drain terminal of first switch  114  is connected to the junction of the anode of the first diode and one end of the primary winding  152  of current transformer  150 . 
   The secondary winding  154  of current transformer  150  is part of the current sensor  160  for measuring the current through the first switch  114 . The secondary winding  154  is connected to a measurement resistor  158  through a fourth diode  156 . The current transformer  150  and the copper trace are typically susceptible to creating a high voltage spike. The current transformer  150  in the embodiment shown in  FIG. 3A  is arranged in the circuit so that the first diode  116  and first capacitor  112  may be disposed much closer to first switch  114  so as to eliminate the need for a resonant snubber for reducing the voltage spike. Preferably the turns ratio of the current transformer is 1:300 or 1:60 typically depending on the current rating. 
   In operation, the current feedback signals (CS+ and CS−) in  FIG. 3  are used to enable current mode control of the first switch  114  using control circuit  142 . The control circuit  142  preferably includes a conventional PWM controller. The first switch  114  is thus preferably operated using voltage mode control for converter  100  and using current mode control for converter  200 . Alternatively, the first switch  114  is operated using conventional voltage mode control.  FIG. 3A  is a preferred embodiment since the current mode control provides improved robustness, in most cases, as compared to simple voltage mode control. The control circuit  144  includes a conventional PWM controller with current mode control preferably provided for second switch  134 . Alternatively, voltage mode control is provided for second switch  134 . Thus, for the power converter according to the preferred embodmient in  FIG. 3A , the first switch  114  and the second switch  134  may be operated using any combination of voltage mode control and current mode control. That is, for power converter  200 , the first switch  114  and the second switch  134  may be operated with one switch operated using voltage mode control and the other using current mode control, with both switches operated using voltage mode control, or with both switches operated using current mode control, depending on the requirements of a particular application. 
   As described above, the present invention achieves the use of lower voltage and Rdson MOSFETs for a boost converter topology for high power applications to enable so as to reduce cost and conduction losses thereby providing increased efficiency. 
   Having disclosed exemplary embodiments, modifications and variations may be made to the disclosed embodiments while remaining within the scope of the invention as described by the following claims.