Abstract:
A circuit having a first circuit portion configured to receive a pulse-width modulated first signal and a second signal, and configured to generate third and fourth signals each responsive to the first and second signals; a second circuit portion configured to receive the third and fourth signals and to generate a fifth signal responsive to both the third and fourth signals; and a third circuit portion configured to control an on/off state of a first switch in response to the fifth signal, wherein the second signal is present at a load path terminal of the first switch. Also, various related methods.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS  
       [0001]     This application claims priority to European patent application no. 06 022 626.3-1233, filed Oct. 30, 2006, hereby incorporated by reference as to it entirety.  
       BACKGROUND  
       [0002]     Switching regulators with half-bridge circuits commonly use a bootstrap circuit having a diode and a bootstrap capacitor connected in series. The bootstrap circuit is typically connected to a node between a high-side transistor and a low-side transistor. To minimize chip size, the size of the bootstrap capacitor is kept small, thereby also keeping the capacitance small. With the reduced capacitance of the bootstrap capacitor, the power consumption of the floating logic being supplied by the bootstrap capacitor becomes more relevant. Ideally, to alleviate the problems associated with a small bootstrap capacitance, the charge stored in the boot strap capacitor should be maximized and the power consumption of the logic circuits supplied by the bootstrap capacitor should be minimized.  
         [0003]     The charge stored in the boot strap capacitor can be increased if the diode connected between the supply voltage and the bootstrap capacitor is replaced by a device having a lower voltage drop than a silicon diode. For this reason it is known to employ a Schottky diode instead of a silicon diode. However, this approach is unfavorable, because when using common chip fabrication processes Schottky diodes cannot be easily integrated and so have to be realized as an external component.  
         [0004]     For various reasons, the high-side transistor is not driven directly by a level-shifted version of an input signal, but rather via a floating latch which is turned on and turned off by short pulses. In order to reduce a degradation of the signal due to noise, low resistor values are used in the level-shifter resulting in a high load of the bootstrap capacitor. The average load of the bootstrap capacitor can be reduced by triggering only short pulses which are then level shifted. This level shifted pulses trigger the floating latch, thus restoring the signal controlling the switching state of the high-side switch.  
         [0005]     There is a general need for a circuit arrangement and for a method for driving a high-side transistor. Such a circuit arrangement may allow for a relatively simple solution which may provide a fail-safe operation without the necessarily using additional refresh circuits, and may further allow integration into a single chip without necessarily using external components.  
       SUMMARY  
       [0006]     Various aspects are described herein. For example, various illustrative circuits are described herein that have a first circuit portion configured to receive a pulse-width modulated first signal and a second signal, and configured to generate third and fourth signals each responsive to the first and second signals; a second circuit portion configured to receive the third and fourth signals and to generate a fifth signal responsive to both the third and fourth signals; and a third circuit portion configured to control an on/off state of a first switch in response to the fifth signal, wherein the second signal is present at a load path terminal of the first switch. Also, various illustrative methods are described herein for performing these and other functions of the circuits.  
         [0007]     These and other aspects will be described with reference to illustrative embodiments in the Detailed Description. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0008]     Various aspects as described herein can be better understood with reference to the following drawings and detailed description. The components in the figures are not necessarily to scale, and instead emphasis is placed upon illustrating the principles of the invention. Moreover, in the figures, like reference numerals designate corresponding parts.  
         [0009]      FIG. 1  is a schematic diagram of a conventional control circuit with a half-bridge comprising a high-side semiconductor switch and a low-side semiconductor switch.  
         [0010]      FIG. 2  is a schematic diagram of an illustrative embodiment of a circuit arrangement comprising a signal conditioning circuit, a floating logic circuit, a floating driver, a high-side semiconductor switch, a bootstrap capacitor and a switching-device for charging the bootstrap capacitor.  
         [0011]      FIG. 3   a  is a schematic diagram of an illustrative embodiment of the floating logic circuit in more detail, comprising a level shifter and a floating latch.  
         [0012]      FIG. 3   b  is an illustrative timing diagram of the signals that may occur in the floating logic circuit shown in  FIG. 3A .  
         [0013]      FIG. 4   a  is a functional block diagram of an illustrative embodiment of the signal conditioning circuit.  
         [0014]      FIG. 4   b  is an illustrative timing diagram of the input and output signals of the single-conditioning circuit shown in  FIG. 4   a.    
         [0015]      FIG. 5   a  is a functional block diagram of another illustrative embodiment of the signal-conditioning circuit.  
         [0016]      FIG. 5   b  is an illustrative timing diagram of the input and output signals of the enhanced signal-conditioning circuit showed in  FIG. 5   a.    
         [0017]      FIG. 6   a  is an illustrative timing diagram of the input signal, the load-current and the phase-feedback signal for continuous conduction mode.  
         [0018]      FIG. 6   b  is an illustrative timing diagram of the input signal, the load-current, the phase-feedback signal and a zero-detection signal for the discontinuous conduction mode.  
         [0019]      FIG. 7   a  is a functional block diagram of another illustrative embodiment of the signal-conditioning circuit using a zero-detection signal.  
         [0020]      FIG. 7   b  is an illustrative timing diagram of the output and the input signals of the signal-conditioning circuit of  FIG. 7   a.   
     
    
     DETAILED DESCRIPTION  
       [0021]     The various aspects described herein may be embodied in various forms. The following description shows by way of illustration various examples in which the aspects may be practiced. It is understood that other examples may be utilized, and that structural and functional modifications may be made, without departing from the scope of the present disclosure.  
         [0022]     Except where explicitly stated otherwise, all references herein to two or more elements being “coupled,” “connected,” and “interconnected” to each other is intended to broadly include both (a) the elements being directly connected to each other, or otherwise in direct communication with each other, without any intervening elements, as well as (b) the elements being indirectly connected to each other, or otherwise in indirect communication with each other, with one or more intervening elements.  
         [0023]     Various illustrative embodiments of a circuit arrangement and a method are described for driving a high-side semiconductor switch, for instance in the output stage of a buck-converter or in switching voltage regulator modules.  
         [0024]     One example of such an arrangement for driving a high-side semiconductor switch comprises a signal-conditioning circuit receiving an input signal, a floating logic circuit connected to the signal-conditioning circuit, and a floating driver circuit connected between the floating logic circuits and a control terminal of the high-side switch. Besides the input signal, which can be pulse-width modulated, the signal conditioning circuit receives a phase-feedback signal from a second load terminal (e.g. a source terminal) of the high-side transistor. By the use of the received signals the signal-conditioning circuit provides a set signal which is set to a first logic level (e.g. a high level) at a first edge (e.g. a raising edge) of the input signal and to a second logic level (e.g. a low level) at a corresponding edge of the phase-feedback signal, but not before a certain delay time has elapsed. The signal conditioning circuit further provides a reset signal, which is set to a first logic level (e.g. a high level) at a second edge (e.g. a falling edge) of the input signal and to a second logic level (e.g. a low level) at a corresponding edge of the phase-feedback signal, but not before the certain delay time has elapsed. All the signals in this signal-conditioning circuit may not be floating but may instead refer to a fixed reference potential, e.g. a ground potential. In essence, the signal conditioning circuit may generate a set pulse at the first edge (e.g. a raising edge) of the input signal and a reset pulse at a second edge (e.g. a falling edge) of the input signal. The width of the pulses may be determined by the time the phase-feed signal needs to react to a set- or a reset pulse. However, a minimum pulse-width may be provided.  
         [0025]     In this example, the floating logic circuits receiving the set signal and the reset signal from the signal-conditioning circuits may provide an output signal, which may be essentially a level-shifted version of the input signal mentioned above. Therefore the set signal showing only short set pulses and the reset signal showing only short reset pulses may be level-shifted to obtain floating set- and reset signals with respect to the potential of the phase-feedback signal, i.e. with respect to the potential of the source terminal of the high-side switch. The floating logic circuit may further comprise a floating latch, which may be triggered by the floating and level-shifted versions of the set- and the reset signal and which may set the above mentioned floating output signal to a first logic level (e.g. a high level) when receiving a set pulse and to a second logic level (e.g. a low level) when receiving a reset pulse.  
         [0026]     The output signal provided by this floating logic circuit may be received by the floating driver circuit which is connected to a control terminal of the high-side switch and which may directly control the switching state of the high-side switch.  
         [0027]     The circuit arrangement may comprise a bootstrap circuit having a bootstrap capacitor with a first terminal connected to the second load terminal (e.g. the source terminal) of the high-side switch and with a second terminal connected to a second supply potential via the load-path of a switching-device, which is also part of the boot strap circuit. The switching-device may comprise a control terminal, which receives a control signal also provided by the above mentioned floating logic circuit. In this example, the floating-logic circuit may be adapted for turning on the switching-device at a first edge of the reset signal (i.e. when receiving the reset pulse) unless it is already switched on and it may be adapted for turning off the switching device at a first edge of the set signal (i.e. when receiving set pulse).  
         [0028]     In another example, the circuit arrangement may further comprise a low-side switch with a first load terminal (e.g. a drain terminal), a second load terminal (e.g. a source terminal) and a control terminal (e.g. a gate terminal), its first load terminal being connected to the second load terminal of the high-side switch and its second load terminal being connected to ground potential. The connected high-side switch and low-side switch together may form a half bridge. The common node of the two switches (the second load terminal of the high-side switch and the first load terminal of the low-side switch respectively) may form the output terminal of the half-bride which is also called the phase terminal providing the phase-feedback signal mentioned above. In this example, the circuit arrangement may also comprise a zero-detection device adapted for directly or indirectly detecting a zero-crossing of a load-current provided by the phase terminal to a load circuit and adapted for providing a zero-detection signal, which is received by the above mentioned signal-conditional circuit.  
         [0029]     When the load current becomes zero while the low-side switch is in an on-state, the low-side switch may switch to a high ohmic state in order to inhibit a discharging of a capacitor in the load circuit through the low-side switch. If the low-side switch has been switched off, the common node of the high-side switch and the low-side switch, i.e. the phase terminal, may become floating and its potential may be determined by the load-circuit. Therefore, the switching device which the boot strapped capacitor is loaded through also may be switched off in order to inhibit discharging of the boot strapped capacitor. This can be done with little effort by generating another reset pulse in the signal conditioning logic which is transported to floating logic via the level-shifter, the floating logic being adapted to turn off the switching-device when receiving a reset signal while the switching device is already in an on-state.  
         [0030]     The charge being stored in the bootstrap capacitor may be maximized by using a switching-device for charging the bootstrap capacitor instead of using a diode with a relatively high voltage drop. The power consumption of the floating logic may be minimized by using the phase-feedback signal which allows short set- and reset pulses, which may be just as long as necessary for triggering the floating gate, thus minimizing the power-consumption of the hole floating logic especially of the level shifter.  
         [0031]     According to an illustrative method of controlling a high-side switch, the high-side semiconductor switch may be controlled such that the high-side switch is turned on at a first edge (e.g. a rising edge) of an input signal and is turned off at a second edge (e.g. a falling edge) of the input signal. The method may comprise, for example, the following: 
        (a) Setting a set signal to first logic level (e.g. a high level) at a first edge (e.g. a rising edge) of an input signal and resetting the set signal to a second logic level (e.g. a low level) at the corresponding edge of a phase-feedback signal but not before a certain delay time has elapsed. The input signal and the set signal both refer to a ground potential; and     (b) Setting a reset signal to a first logic level (e.g. a high level) at a second edge (e.g. a falling edge) of the input signal and resetting the reset signal to a second logic level (e.g. a low level) at a corresponding edge of a phase-feedback signal but not before a certain delay time has elapsed.        
 
         [0034]     Like the set signal, the reset signal may also refer to ground potential. In the steps (a) and (b), set- and reset pulses are generated in response to first and second edges of the input signal respectively. The duration of the set- and the reset pulses may be just as long as necessary for the phase-feedback signal to react. The illustrative method may further include the following: 
        (c) Level shifting the set signal and the reset signal to provide floating versions thereof which refer to the potential of the phase-feedback signal; and     (d) Setting an output signal to a first logic level (e.g. a high level) at a second edge (e.g. a falling edge) of the floating version of the set signal and resetting the output signal to a second logic level (e.g. a low level) at the corresponding edge of the floating version of the reset signal.        
 
         [0037]     Like the set- and the reset signals, the output signal may also be floating and refer to a potential of the phase-feedback signal. The illustrative method may further include: 
        (e) Turning on or off the high-side switch dependent on the logic level of the output signal.        
 
         [0039]     According to another example, the method may additionally include: 
        (f) Turning on a switching-device for charging a bootstrap capacitor at a first edge (e.g. a rising edge) of the reset signal; and     (g) Turning off the switching device at a first edge (e.g. a rising edge) of the set signal.        
 
         [0042]     According to a further example, the method may include: 
        (h) Detecting a zero-crossing of a load-current provided by a half-bridge formed of the high-side switch and a low-side switch, setting a zero-detection signal to a first logic level (e.g. a high level) when a zero-crossing has been detected and resetting it to a second logic level (e.g. a low level) after a certain delay time has elapsed;     (i) Turning off the above-mentioned switching-device at a first edge of zero-detection signal; and     (j) Turning off the low-side switch at a first edge of the zero detection signal.        
 
         [0046]     A conventional switching regulator with a half-bridge circuit is shown in  FIG. 1 . The half-bridge circuit of  FIG. 1  has two n-channel metal-oxide-semiconductor (MOS) transistors T 1  and T 2  whose drain-source paths are connected in series. This series circuit is connected to a supply potential VCC and to a reference potential GND. A control circuit  1  is employed for driving the MOS transistors T 1  and T 2 . The control circuit is connected to the gates of the MOS transistors T 1  and T 2 , to the supply potential VCC, as well as to the reference potential GND for supply purposes. A bootstrap-supply circuit provides a bootstrap-supply voltage VBTS to the control circuit  1  in order to enable the control circuit  1  to generate a driving signal with a potential high enough to drive the gate of the high-side transistor T 1  of the half-bridge to.  
         [0047]     A load circuit  3  having a complex impedance is connected between a common node of the high-side transistor T 1  and the low-side transistor T 2  and ground potential GND. The load circuit  3  may comprise an inductor connected in series to a capacitor and a resistor connected in parallel to the capacitor. The voltage drop across the resistor, i.e. the output voltage, is substantially determined by an input signal PWM.  
         [0048]     The control circuit  1  controls the gates of the high-side transistor T 1  and the low-side transistor T 2  such that the potential of the common node substantially follows the input signal PWM.  
         [0049]     The circuit of  FIG. 1  comprises a commonly used bootstrap circuit having a diode and a bootstrap capacitor C connected in series between the supply potential VCC and the source terminal of the high-side transistor T 1 , i.e. the common node of the high-side transistor T 1  and the low-side transistor T 2 . If the common node of the transistors goes to a low level, the bootstrap capacitor C is charged through the diode up to a voltage equal to the supply potential VCC minus the voltage drop across the diode and across the low side transistor T 2 . If the common node of the transistors T 1  and T 2  switches to a high level, discharging of the bootstrap capacitor C is inhibited by the diodes and consequently the bootstrap capacitor C can provide a floating bootstrap supply voltage VBTS with respect to the source terminal of the high-side transistor T 1 . The control circuit  1  is connected to the common node of the diode and the bootstrap capacitor C, thus enabling the control circuit  1  to provide signals with a potential well above the potential of the source terminal of the high-side transistor T 1 .  
         [0050]      FIG. 2  shows a schematic of an illustrative embodiment of a circuit arrangement that comprises a signal-conditioning circuit  12 , a floating logic circuit  13 , a floating driver  14  for driving a high side semiconductor switch T 1  and a driver  15  for driving a low side semiconductor switch T 2 . The high side switch T 1  and the low side switch T 2  are connected in series to form a half bridge. Therefore, the drain terminal of the high side switch T 1  receives a first supply voltage Vin, the source terminal of the high side switch T 1  and drain terminal of the low side switch T 2  are connected together to form a phase terminal PHS, and the source terminal of the low side switch T 2  is connected to ground potential. The common node of the high side switch T 1  and the low side switch T 2 , i.e. the phase terminal PHS, is connected to a load circuit  30 .  
         [0051]     The signal conditioning circuit  12  receives an input signal S 121  and a phase-feedback signal S 122  and provides a set signal S 131  and a reset signal S 132 . The input signal S 121  is a pulse width modulated signal whose duty-cycle determines the mean output voltage of the phase terminal PHS with respect to a ground potential (i.e. the output voltage of the half bridge). The signal conditioning circuit  12  is adapted for providing a set signal S 131  with set pulses and a reset signal S 132  with reset pulses. At a rising edge of the input signal S 121  the set signal S 131  is set from a low level to a high level, and at a corresponding edge of the phase-feedback signal (i.e. the output of the half bridge) S 122  the set signal is reset to a low level after a certain delay time TD has elapsed. At a falling edge of the input signal S 121  the reset signal S 132  is set from a low level to a high level, and at a corresponding edge of the phase-feedback signal S 122  the reset signal S 132  is reset to low level after the certain delay time TD has elapsed. Substantially the set- and the reset signal may be in the form of set pulses and reset pulses triggered by rising and falling edges of the input signal S 121  respectively. The duration of each set pulse (or reset pulse) is determined by the time between a rising edge (or a falling edge) of the input signal S 121  and a corresponding edge of the phase-feedback signal S 122 . To allow for the width of the set- and the reset pulses to be big enough to reliably trigger a latch receiving the set- and the reset signals, a minimum pulse width of the set- and the reset pulses is provided by waiting at least the certain delay time TD before resetting the set or the reset signal from a high level to a low level.  
         [0052]     The floating logic circuit  13  receives the set signal S 131  and the reset signal S 132  and provides on the one hand a floating output signal S 141  for controlling the switching state of the high side switch T 1  and on the other side a control signal SM 3  for controlling the charging process of a bootstrap capacitor C by means of another semiconductor switch M 3  (explained below).  
         [0053]     The output signal S 141  is received by a floating driver  14  which is directly connected to a gate terminal of the high side switch T 1  which is realized as an n-channel MOSFET. The floating control signal SM 3  is connected to the gate of another semiconductor switch which is realized as a p-channel MOSFET connected between a second power supply terminal Vcc and a first terminal of the bootstrap capacitor C. The first terminal of the bootstrap capacitor C provides a floating bootstrap supply voltage Vbts with respect to the potential of the phase terminal PHS. The second terminal of the bootstrap capacitor C is connected to the source terminal of the high side switch T 1 , i.e. the phase terminal PHS. The internals of the floating logic circuit  13  will be explained later with reference to  FIGS. 3   a  and  3   b.    
         [0054]     The input signal S 121  is provided by a shoot-through protection-logic  11  which is adapted for protecting the half bridge from cross-conduction. The input signal S 121  is substantially a time delayed version of a master signal PWM received by the shoot-through protection-logic  11 . The shoot-through protection-logic  11  also provides another input signal S 151  for driving the low side switch T 2  via driver  15 . The input signal S 151  is substantially an inverted version of the master signal PWM. The short time delay between the input signal S 121  and the other input signal S 151  provides that the high side switch T 1  is not opened before the low side switch T 2  is closed.  
         [0055]      FIG. 3   a  shows an illustrative embodiment of the floating logic circuit  13  in more detail. A level shifter  131  receives the set signal S 131  and the reset signal S 132  and provides inverted and level shifted versions (S 131 ′ and S 132 ′) thereof. The level shifted and floating versions of the set- and the reset signal (S 131 ′, S 132 ′) are received by a floating latch  132 . The floating latch  132  provides an output signal S 141  and a control signal SM 3  for controlling the charging of the bootstrap capacitor. The function of the floating latch  132  is explained below by means of timing diagrams shown in  FIG. 3   b.    
         [0056]     The level shifter  131  comprises a first transistor M 1  with a source terminal connected to ground potential, a gate terminal receiving the set signal S 131  and a drain terminal connected to the bootstrap supply voltage VBTS via a parallel circuit of a resistor R 1  and a zener-diode D 1 . The drain terminal of the first transistor M 1  provides the level-shifted and floating version S 131 ′ of the set signal. If the set signal S 131  received by the gate of the first transistor M 1  is on a low level, the transistor M 1  assumes a high ohmic state and the drain terminal of the first transistor M 1  is pulled up to the bootstrap voltage VBTS by the resistor R 1 . Consequently the level shifted version S 131 ′ of the set signal is at a high level with a voltage VBTS over ground potential GND. When the set signal S 131  changes to a high level, the first transistor M 1  is switched on and the drain of the first transistor M 1  is pulled down to a voltage equal to the bootstrap voltage VBTS minus the zener-voltage Vz determined by the zener-diode D 1 . The zener-voltage Vz of the diode D 1  is approximately as large as the voltage drop over the bootstrap capacitor C. In addition to its level shifting function, the level shifter  131  acts as an inverter. Inverting the set- and the reset signals is not compulsory but may allow for a simpler configuration of the level-shifter  131 . To compensate for this inversion, the following floating latch  132  is triggered by low level pulses instead of high level pulses or falling edges instead of rising edges. The level-shifting for the reset signal S 132  is completely analogous to the level shifting for the set signal S 131 . Therefore, a second transistor M 2 , a second resistor R 1  and a second zener-diode D 2  are employed.  
         [0057]     The function of the floating logic circuit  13  can be understood easier on the basis of the illustrative timing diagrams shown in  FIG. 3   b . The first diagram shows one possible input signal S 121 . Here the duty-cycle of the pulse width modulated input signal is 50 percent. The set signal S 131  has a short set pulse at each rising edge of the input signal S 121 . The reset signal S 132  has a reset pulse at each falling edge of the input signal S 121 . The fourth and the fifth diagram in  FIG. 3   b  show the level-shifted versions of the set- and the reset signal (S 131 ′, S 132 ′). The level shifted versions of the set- and reset signals are floating with respect to the phase terminal PHS, i.e. the source terminal of the high-side switch. The floating latch  132  sets the output signal S 141  to a high level at a falling edge of the level-shifted version of the set signal S 131 ′ and resets the output signal S 141  to a low level at a falling edge of the level-shifted version of the reset signal S 132 ′.  
         [0058]     In essence the output signal S 141  in this example follows the input signal S 121 , with the only difference that the output signal S 141  is floating and does not refer to ground potential (but to the potential of the phase terminal PHS). The control signal SM 3  is set to a high level when a reset pulse is received by the floating latch  132  and set to a low level when a set pulse is received by the floating latch  132 . In this case the control signal SM 3  is substantially an inverted version of the output signal S 141 .  
         [0059]     When the output signal S 141  is at a low level, the high side switch is opened and the potential of the source terminal of the high side switch T 1  falls close to ground potential GND. In this state the bootstrap capacitor C can be charged and the p-channel MOSFET M 3  has to be switched to an on-state in order to charge the bootstrap capacitor C. When the output signal goes to a high level the high side switch T 1  is switched on and the potential of the source terminal of the high side switch goes up close to the supply voltage Vin. In that case the control signal SM 3  has to change to a low level in order to switch off the p-channel MOSFET M 3  inhibiting the discharging of the bootstrap capacitor C.  
         [0060]      FIG. 4   a  shows a simplified illustrative schematic of the signal conditioning circuit  12 , and  FIG. 4   b  shows illustrative corresponding timing diagrams. It is noted that the implementation shown in  FIG. 4   a  is just an example; there are many ways for realizing the same boolean function. The input signal S 121  and the phase-feedback signal S 122  are linked together to provide the set signal S 131  by the use of a NOT-gate  120   a  and an AND-gate  121   a  and to provide the reset signal S 132  by the use of a NOT-gate  120   b  and an AND-gate  121   b . The set signal S 131  and the reset signal S 132  obey the following boolean relation: 
 S131=S121 &amp;  S122   (1).  S132=  S121  &amp; S122  (2).  
         [0061]     The effect of the logic circuit shown in  FIG. 4   a  on the signals involved can be seen in the timing diagrams shown in  FIG. 4   b . The first diagram again shows the input signal S 121  with a duty-cycle of 50 percent. At a rising edge of the input signal S 121  the set signal S 131  is set to a high level and is not reset to a low level before a rising edge in the phase-feedback signal S 122  has occurred. If the phase-feedback signal S 122  erroneously falls back to a low level during a high level of the input signal S 121  another set pulse is triggered due to the phase-feedback. The phase-feedback signal S 122  is analogously applied to the reset signal S 132 . The reset signal S 132  is set to a high level at a falling edge of the input signal S 121  and is not set back to a low level before a falling edge in the phase-feedback signal S 122  has occurred. This use of the phase-feedback may allow set- and reset pulses with a minimal pulse width and may allow the automatic generation of additional set or reset pulses if the phase-feedback signal S 122  indicates an erroneous level change due to a glitch which can trigger the floating latch in error.  
         [0062]      FIG. 5   a  shows an illustrative modified version of signal conditioning circuit  12 , and  FIG. 5   b  shows illustrative corresponding time diagrams. The only difference between the circuit shown in  FIG. 4   a  and the circuit shown in  FIG. 5   a  are the monostable multivibrators  122   a  and  122   b  and the OR-gates  123   a  and  123   b . The monostable multivibrator  122   a  is triggered by a rising edge of the input signal S 121  and its output X is combined with the output of the end-gate  121   a  by an or-gate  123   a . The output of the OR-gate  123   a  yields the set signal S 131 . The other monostable multivibrator  122   b  is triggered by a falling edge of the input signal S 121  and its output Y is combined with the output of the end-gate  121   b  by the OR-gate  123   b . The output of the OR-gate  123   b  yields the reset signal S 132 . The set signal S 131  and the reset signal S 132  obey the following equations: 
 S131=(S121 &amp;  S122 )OR X  (3).  S132=(  S121  &amp; S122)OR Y  (4).  
         [0063]     The function of the signal conditioning circuit  12  from  FIG. 5   a  is easily understood with the help of the timing diagrams shown in  FIG. 5   b . The first diagram shows an illustrative input signal S 121  which is a pulse-width modulated signal with a duty-cycle of 50 percent. Like in the previous circuit of  FIG. 4   a , the set signal S 131  is set from a low level to a high level at a rising edge of the input signal S 121 , but it is not reset to a low level unless a certain time delay TD, which depends on the internals of the monostable multivibrator  120   a , has elapsed even if the phase-feedback signal S 122  has reacted and changed to a high level in the meantime. Only if the phase-feedback signal S 122  erroneously does not change to a high level within the delay time TD the set signal S 131  is not reset to a low level before the desired level change in the phase-feedback signal S 122  has observed. The reset signal S 132  is set from a low level to a high level at a falling edge of the input signal S 121  and set back at a corresponding falling edge of the phase-feedback signal S 122  but—like the set signal S 131 —is not set back to a low level before a certain time delay TD has elapsed.  
         [0064]     The circuit shown in  FIG. 5   a  behaves essentially like the circuit shown in  FIG. 4   a , with the only difference being that with the circuit shown in  FIG. 5   a , a minimum pulse width of the set and the reset pulses is provided.  
         [0065]     The circuit arrangement as explained by now may work well as long as the circuit remains in a continuous conduction mode, which would mean that the output current IL provided by the half bridge (T 1 , T 2 ) would not become zero. The continuous conduction mode is illustrated by way of example on the basis of timing diagrams in  FIG. 6   a . The first timing diagram shows an illustrative input signal S 121  with a duty-cycle of approximately 30 percent. The second timing diagram shows the resulting output current IL in continuous conduction mode. The resulting phase-feedback signal S 122  is shown in the third timing diagram of  FIG. 6   a . The phase-feedback signal S 122  substantially follows the input signal S 121 . The slight slope in the phase-feedback signal S 122  is due to the voltage drop over the on-resistance Ron of the high side switch T 1  or over the low side switch T 2  dependent on the logic level of the phase-feedback signal S 122 . During an on-cycle of the input signal S 122  the phase-feedback signal S 122  is given by 
 
 S 122= V in− IL·R on,  (5). 
 
         [0066]     During an off-cycle of the input signal S 122  the phase-feedback signal S 122  is given by 
 
 S 122 =IL·R on.  (6) 
 
         [0067]      FIG. 6   b  shows illustrative timing diagrams valid for the circuit arrangement operating in a discontinuous conduction mode. The first timing diagram again shows an illustrative input signal S 121 . The resulting output current IL in discontinuous conduction mode does—in contrast to continuous conduction mode—become zero during an off-cycle of the input signal S 121 . Without any additional provisions the current would become negative, i.e. the current would flow from the load circuit  30  into the half bridge (T 1 , T 2 ). This behavior (a negative load current IL) may be undesired, because a capacitor in the load circuit or even the bootstrap capacitor C could be discharged. In order to avoid an inverse load current IL a zero crossing of the load current IL is detected and upon a zero crossing of the load current IL the low side switch T 2  is switched off, thus inhibiting a reverse output current and making the phase terminal PHS of the half bridge floating. The voltage of the phase terminal PHS is then determined by capacitances in the load circuit  30 .  
         [0068]     A zero detection device may be provided in an enhanced embodiment for detecting the above-mentioned zero crossing of the output current IL. Such a zero detection device may be easily implemented by using a comparator which detects the phase-feedback signal S 122  becoming zero. From equation (5) and (6), as well as from  FIG. 6   b , it is clear that a phase-feedback signal S 122  of zero is equivalent to a load-current IL of zero.  
         [0069]     The zero detection device provides a zero-detection signal S 123  which indicates a zero-crossing of the output current IL (or of the phase-feedback signal S 122 ) by means of a short pulse with a rising edge at the time the zero crossing occurs.  
         [0070]     As explained above, the low side switch T 2  may be switched off in order to inhibit a negative load current IL. This may not be enough to inhibit a discharge of the bootstrap capacitor C. Therefore, the p-channel MOSFET M 3  controlling the charging of the bootstrap capacitor C may be switched off when a zero-crossing of the load current IL is detected. Since the control signal SM 3  controlling the switching state of the p-channel MOSFET M 3  is also a floating signal with respect to the potential of the phase terminal, the p-channel MOSFET M 3  is not switched off by the zero-crossing detection device directly. This may be done via the floating logic circuit  13 . Therefore, another modified signal conditioning circuit  12  may be used to transmit the zero-detection signal S 123  to the floating logic circuit  13 . An example of such an embodiment of the signal-conditioning circuit  12  is shown in  FIG. 7   a . The circuit is substantially like the circuit shown in  FIG. 5   a , with the only difference that another monostable multivibrator  122   c  and a three input OR-gate  123   b ′ is used to trigger an additional reset pulse at a rising edge in the zero-detection signal S 123 . More precisely, a rising edge in the zero detection signal S 123  triggers a short pulse by the use of the monostable multivibrator  122   c . This signal Z is combined with the output of the AND-gate  121   b  and the monostable multivibrator  122   b  by the use of the three-input OR-gate  123   b ′. The output of this OR-gate  123   b ′ yields the reset signal S 132 , as follows: 
 
S132=(  S121  &amp; S122)OR Y OR Z  (7). 
 
         [0071]     This additional reset pulse triggered by the zero-detection signal S 123  is transmitted to the floating latch  132  through the level shifter  131 . In this embodiment of the invention the floating latch  132  is adapted such, that the control signal SM 3  is set to a low level upon an additional reset pulse while the p-channel MOSFET M 3  is in an on-state. This behavior can be easily understood with the help of  FIG. 7   b.    
         [0072]      FIG. 7   b  shows illustrative timing diagrams of the input signal S 121 , the reset signal S 132 , the floating version of the reset signal S 132 ′, the output current IL, the zero-detection signal S 123 , and the control signal SM 3  controlling the charging of the bootstrap capacitor C. The first timing diagram again shows an illustrative input signal S 121  with a duty-cycle of 50%. At the falling edge of the input signal S 121  a reset pulse is triggered in the reset signal S 132  which is shown in the second timing diagram of  FIG. 7   b . The third timing diagram shows the floating and level-shifted version S 132 ′ of the set signal and the fourth timing diagram shows the resulting output-current IL. If the output-current IL becomes zero, a pulse in the zero-detection signal S 123  is triggered and—due to the modified signal conditioning circuit  12  shown in  FIG. 6   a —this pulse is “copied” into the reset signal S 132  and consequently also into the floating and level-shifted version S 132 ′. This additional reset pulse, which is due to the zero-detection in the output current IL, triggers the floating latch  123  setting the control signal SM 3  to a low level, thus switching off the p-channel MOSFET M 3  and thus stopping the charging of the bootstrap capacitor C.  
         [0073]     The logic levels used for all the illustrative signals described herein depend on the actual implementation of the logic circuit and could be inverted depending on the actual implementation of the logic circuit.