Abstract:
A capacitive sensing device includes an antenna electrode for emitting an alternating electric field in response to an alternating voltage caused in the antenna electrode and a control and evaluation circuit that includes a transimpedance amplifier configured to maintain the alternating voltage equal to an alternating reference voltage on a reference voltage node by injecting a current into the antenna electrode and to measure the current. The control and evaluation circuit includes a microcontroller and a multiplexer configured and arranged to switch the antenna electrode alternately to a current input of the transimpedance amplifier and to the alternating reference voltage node. The microcontroller is configured to control the multiplexer. The multiplexer and the transimpedance amplifier and a low-pass filter operatively connected to the transimpedance amplifier form together a synchronous rectifier arrangement. The current input node of the transimpedance amplifier is AC-coupled to the reference voltage node by a protection capacitor.

Description:
TECHNICAL FIELD 
     The present invention generally relates to a capacitive sensing device, e.g. for detecting the absence or presence of an occupant seated on a vehicle seat. 
     BACKGROUND OF THE INVENTION 
     A capacitive sensor or capacitive sensing device, called by some electric field sensor or proximity sensor, designates a sensor, which generates a signal responsive to the influence of what is being sensed (a person, a part of a person&#39;s body, a pet, an object, etc.) upon an electric field. A capacitive sensor generally comprises at least one antenna electrode, to which is applied an oscillating electric signal and which thereupon emits an electric field into a region of space proximate to the antenna electrode, while the sensor is operating. The sensor comprises at least one sensing electrode at which the influence of an object or living being on the electric field is detected. In some (so-called “loading mode) capacitive occupancy sensors, the one or more antenna electrodes serve at the same time as sensing electrodes. In this case, the measurement circuit determines the current flowing into the one or more antenna electrodes in response to an oscillating voltage being applied to them. The relationship of voltage to current yields the complex impedance between the one or more antenna electrodes and ground. In an alternative version of capacitive sensors (“coupling mode” capacitive sensors), the transmitting antenna electrode(s) and the sensing electrode(s) are separate from one another. In this case, the measurement circuit determines the current or voltage that is induced in the sensing electrode when the transmitting antenna electrode is operating. 
     The different capacitive sensing mechanisms are explained in the technical paper entitled “Electric Field Sensing for Graphical Interfaces” by J. R. Smith, published in Computer Graphics I/O Devices, Issue May/June 1998, pp 54-60. The paper describes the concept of electric field sensing as used for making non-contact three-dimensional position measurements, and more particularly for sensing the position of a human hand for purposes of providing three-dimensional positional inputs to a computer. Within the general concept of capacitive sensing, the author distinguishes between distinct mechanisms he refers to as “loading mode”, “shunt mode”, and “transmit mode” which correspond to various possible electric current pathways. In the “loading mode”, an oscillating voltage signal is applied to a transmit electrode, which builds up an oscillating electric field to ground. The object to be sensed modifies the capacitance between the transmit electrode and ground. In the “shunt mode”, an oscillating voltage signal is applied to the transmit electrode, building up an electric field to a receive electrode, and the displacement current induced at the receive electrode is measured, whereby the displacement current may be modified by the body being sensed. In the “transmit mode”, the transmit electrode is put in contact with the user&#39;s body, which then becomes a transmitter relative to a receiver, either by direct electrical connection or via capacitive coupling. “Shunt mode” is alternatively referred to as the above-mentioned “coupling mode”. 
     Capacitive occupant sensing systems have been proposed in great variety, e.g. for controlling the deployment of one or more airbags, such as e.g. a driver airbag, a passenger airbag and/or a side airbag. U.S. Pat. No. 6,161,070, to Jinno et al., relates to a passenger detection system including a single antenna electrode mounted on a surface of a passenger seat in an automobile. An oscillator applies on oscillating voltage signal to the antenna electrode, whereby a minute electric field is produced around the antenna electrode. Jinno proposes detecting the presence or absence of a passenger in the seat based on the amplitude and the phase of the current flowing to the antenna electrode. 
     U.S. Pat. No. 6,392,542, to Stanley, teaches an electric field sensor comprising an electrode mountable within a seat and operatively coupled to a sensing circuit, which applies to the electrode an oscillating or pulsed signal having a frequency “at most weakly responsive” to wetness of the seat. Stanley proposes to measure phase and amplitude of the current flowing to the electrode to detect an occupied or an empty seat and to compensate for seat wetness. 
     Others had the idea of using the heating element of a seat heater as an antenna electrode of a capacitive occupancy sensing system. WO 92/17344 A1 discloses a an electrically heated vehicle seat with a conductor, which can be heated by the passage of electrical current, located in the seating surface, wherein the conductor also forms one electrode of a two-electrode seat occupancy sensor. 
     WO 95/13204 discloses a similar system, in which the oscillation frequency of an oscillator connected to the heating element is measured to derive the occupancy state of the vehicle seat. More elaborate combinations of a seat heater and a capacitive sensor are disclosed, for instance, in U.S. Pat. No. 7,521,940, US 2009/0295199 and U.S. Pat. No. 6,703,845. 
     BRIEF SUMMARY 
     The present invention provides a capacitive sensing device (which may be combined or not with a heating element), which can be manufactured at a competitive price. 
     In accordance with the invention, a capacitive sensing device comprises an antenna electrode for emitting an alternating electric field in response to an alternating voltage caused in the antenna electrode and a control and evaluation circuit that comprises a transimpedance amplifier configured to maintain the alternating voltage equal to an alternating reference voltage on a reference voltage node by injecting a current into the antenna electrode and to measure the current. The control and evaluation circuit comprises a microcontroller and a multiplexer configured and arranged to switch said antenna electrode alternately to a current input of the transimpedance amplifier and to the alternating reference voltage node, the microcontroller being configured to control the multiplexer with a digital control signal. The multiplexer and the transimpedance amplifier and a low-pass filter operatively connected to the transimpedance amplifier form together a synchronous rectifier arrangement. The current input node of the transimpedance amplifier is AC-coupled to the reference voltage node by a protection capacitor. As will be appreciated, the protection capacitor provides a low-impedance path between the current input node and the transimpedance amplifier and thus substantially prevents high-frequency currents from distorting the useful signal components output by the synchronous rectifier. Preferably, a second protection capacitor AC-couples the reference voltage node to circuit ground. 
     As will be appreciated, the protection capacitor guarantees that the input of the transimpedance amplifier is always connected to the reference voltage node via a comparatively small AC impedance, irrespective of the position of the multiplexer. Keeping the input AC impedance of the measurement electronics low implies that substantially all high-frequency currents (i.e. with frequencies well above the frequency of the alternating reference voltage) induced in the antenna electrode flow also through the measurement path starting with the multiplexer, instead of flowing directly into the reference voltage node through the parasitic impedances between the antenna electrode and the reference voltage node. Since the demodulation is effected by the multiplexer, the useful information is comprised in the DC component of the current that flows into the transimpedance amplifier. Furthermore, the DC component of that current cannot flow away across the protection capacitor. Accordingly, while the protection capacitor deviates alternating current to the reference voltage node, it does not reduce the sensitivity of the measurement electronics regarding the impedance to be measured. 
     In accordance with a preferred embodiment of the invention, the microcontroller comprises a measurement input and a digital output, the measurement input being operatively connected to the synchronous rectifier arrangement for receiving an output signal thereof as a measurement input signal, and the digital output being configured to provide a digital signal. A further low-pass filter is operatively connected to the digital output for generating the alternating reference voltage by low-pass-filtering the digital signal. As will be appreciated, this embodiment of the invention takes advantage from the presence of a microcontroller within the control and evaluation circuit for the generation of the alternating reference voltage. Since nowadays&#39; capacitive sensors anyway typically comprise a microcontroller for processing the measurements and/or communicating with other devices, low-pass-filtering a digital signal for generating the substantially sinusoidal alternating reference voltage represents a very cost-efficient alternative to prior solutions (typically using active components, such as e.g. an oscillator or circuits with transistors and operational amplifiers). 
     Preferably, the capacitive sensing device comprises a driven shield electrode operatively connected (e.g. to the low-pass filter that produces the alternating reference voltage) for having the alternating reference voltage applied to. A driven shield electrode may be used, in particular, in applications where the antenna electrode is arranged relatively close to a grounded surface. In this case, a driven shield electrode may be arranged between the antenna electrode and the grounded surface, thus reducing the capacitance between the antenna electrode and the grounded surface and making the capacitive sensing device more sensitive to smaller changes of that capacitance, which are e.g. caused by the proximity of a person&#39;s hand or body. 
     If the antenna electrode is arranged at a certain distance from the control and evaluation circuit, the capacitive sensing device preferably comprises a shielded cable, including a core conductor and a shield conductor surrounding the core conductor, with the antenna electrode being operatively connected with the control and evaluation circuit via the core conductor and with the shield conductor being operatively connected (e.g. to the driven shield electrode and/or to the reference voltage node) so as to receive the alternating reference voltage. The shield conductor prevents the wire between the antenna electrode and the control and evaluation circuit to capacitively couple to ground, which would otherwise induce an undesirable measurement offset that depends on the length of the wire. 
     The low-pass filter operatively connected to the digital output preferably comprises an LC low-pass filter. 
     According to a preferred embodiment of the invention, the capacitive sensing device comprises at least one normative impedance controlledly switchable in parallel to the antenna electrode with a switch arranged in series with the at least one normative impedance and controlled by the microcontroller. 
     The transimpedance amplifier preferably has a current input operatively connected to the antenna electrode, a reference voltage input operatively connected to said reference voltage node for receiving the alternating reference voltage and an output operatively connected with the current input via a feedback network. The transimpedance amplifier is preferably configured for preventing a difference in electrical potential between the current input and the reference voltage input by causing a current to flow across the reference network and into the antenna electrode, the output being configured to output a measurement voltage indicative of the current. 
     The low-pass filter of the synchronous rectifier arrangement preferably includes an offset correction circuit. 
     Preferably, the multiplexer is configured to switch the antenna electrode to the current input during a first half of a period of the alternating reference voltage and to the reference voltage input during a second half of the alternating reference voltage. 
     According to a preferred embodiment of the invention, a first shunt capacitor is connected between the antenna electrode and the reference voltage input of the transimpedance amplifier and a second shunt capacitor is connected between the reference voltage node and circuit ground. The second shunt capacitor in this case corresponds to the above-mentioned second protection capacitor. As will be appreciated, the first and second shunt capacitors provide for a path from the antenna electrode to circuit ground that bypasses the transimpedance amplifier and the microcontroller. That path has low impedance for high-frequency current components of disturbance or interference currents that may be injected into the antenna electrode. 
     Capacitive sensing devices in accordance with the invention may be used, for example, in a capacitive trunk opener (which opens the trunk of a car if an approaching leg is detected), in a capacitive door opener, or in integrated capacitive sensing and heating systems, e.g. in a vehicle seat or a steering wheel. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Further details and advantages of the present invention will be apparent from the following detailed description of a not limiting embodiment with reference to the attached drawing, wherein: 
         FIG. 1  is a block schematic diagram of a capacitive sensing device in accordance with the preferred aspects of the invention. 
     
    
    
     DESCRIPTION OF PREFERRED EMBODIMENTS 
       FIG. 1  shows an embodiment of a capacitive sensing device  10  in accordance with the preferred aspects of the invention. The capacitive sensing device  10  comprises a conductive antenna electrode (“sense electrode”)  12  and a conductive driven shield electrode (“guard electrode”)  14 . Sense and guard electrodes  12 ,  14  are electrically insulated from each other. The a priori unknown, complex impedance  16  between the sense electrode  12  and ground is the impedance to be measured by the capacitive sensing device  10 . Sense electrode  12  and guard electrode  14  are connected to a control and evaluation circuit (“measurement electronics”)  18  by a shielded cable  20 . 
     The control and evaluation circuit  18  comprises a conductor, hereinafter referred to as guard node  22 , to which a sine voltage with a frequency comprised in the range of 100 kHz to 500 kHz is applied. It shall be noted, however, that variants of the present embodiment may employ other frequencies in the range from 10 kHz to 10 MHz without departing from the scope of the invention. The sine voltage on the guard node  22  is also called the guard voltage or the guard sine wave and represents the above-mentioned alternating reference voltage. The guard node  22  is connected with the guard electrode  14  by the shield conductor  24  of the shielded cable  20 . 
     The sine voltage on the guard node  22  is generated by a low-pass LC filter  26 . The guard node  22  corresponds to the output of the low-pass LC filter  26 . The input of the low-pass LC filter  26  is connected with a PWM (pulse-width modulation) output  28  of a microcontroller  30 , which, during operation, applies a PWM-digital signal (a square wave) with the desired frequency to the low-pass LC filter  26 . The low-pass LC filter  26  converts the digital signal into the sinusoidal guard voltage basically by letting pass only the fundamental frequency of the digital signal and suppressing the harmonics thereof. 
     In the illustrated embodiment, the low-pass LC filter  26  comprises of source terminating resistance  32 , a first capacitor  34 , an inductor  36 , a second capacitor  38 , load termination resistance  40  and load termination decoupling capacitor  42 . Capacitor  42  ensures that the DC component of the output voltage is one half of the operating voltage of microcontroller  30 . Preferably, the cutoff frequency of the low-pass LC filter  26  is chosen such that it lies above the operating frequency range of the guard voltage. However, if some attenuation of the fundamental frequency is tolerable, the cutoff frequency of the low-pass LC filter  26  could also lie slightly below the fundamental frequency, which may lead to a better suppression of the harmonics relative to the fundamental frequency. Preferably, filter topology and filter type are chosen such that for the given cutoff frequency, capacitance  38  is maximized. The purpose of this capacitance maximization is detailed later. In the illustrated embodiment, a 3-pole, 0.1 dB ripple Chebychev low-pass filter is used. This choice represents a good compromise between low component count leading to low cost, harmonics suppression and maximization of the capacitance  38 . Obviously, other filter topologies can lead to similar results. A noticeable advantage of using a digital signal from a microcontroller in combination with an LC filter for generating the guard voltage is low overall cost. Except for the microcontroller  30 , no active devices, like amplifiers consisting of transistors and/or operational amplifiers, are employed. Since a microcontroller is anyway required in the measurement system, its provision does not involve any extra costs. 
     The sense electrode  12  is connected with a so-called sense node  44  of the control and evaluation circuit  18  via the core conductor  46  of the shielded cable  20 . The sense node  46  is the input node of the measurement electronics and the beginning of the measurement path. The measurement path of the control and evaluation circuit  18  includes, as major components, a multiplexer  48 , a transimpedance amplifier  50  and a low-pass filter with a (fixed) gain and an offset correction  52 . 
     The sense node  44  is connected to the common node of SPDT (single pole, double throw) multiplexer  48 . The multiplexer  48  alternatingly connects the sense node  44  to the guard node  22  during one half of the period of the guard voltage and to the current input  54  of the transimpedance amplifier  50  during the other half of the period of the guard voltage. The transimpedance amplifier  50  has a current input, which, when left open, has a defined voltage equal to the alternating reference voltage. The switching action of the multiplexer  48  is controlled by a second PWM output  57  of the microcontroller  30 . The frequencies of the signals on the PWM outputs  28  and  57  are identical and the software of the microcontroller controls the phase shift between these signals. The microcontroller  30  alternates the phase shift between two phase shift values, which are preferably 90 degrees apart. Measuring the current flowing into the sense node  44  for each of these phase shift values yields two measures, from which the complex impedance or the complex conductance between the sense electrode  12  and ground may be calculated. 
     The transimpedance amplifier  50  comprises a first capacitor  56  connected between the current input  54  of the transimpedance amplifier  50  and the guard node  22 . This capacitance guarantees that the sense node  44  is always connected to the guard node via a comparatively small AC impedance, irrespective of the position of the multiplexer switch  48 . Keeping the input AC impedance of the measurement electronics low between the sense node  44  and the guard node  22  compared to the parasitic impedance between the sense electrode  12  and the guard electrode  14 , the impedance of the cable  20  and the impedance of EMI (electromagnetic interference) protection capacitor  58 , implies that substantially all the current flowing through the unknown impedance  16  flows also through the measurement path starting with the multiplexer  48 , instead of flowing into the guard node  22  through the parasitic impedances mentioned above. In this context, it is worthwhile noting that, since the demodulation is effected by multiplexer  48 , the useful information is comprised in the DC component of the current that flows into the current input  54  of the transimpedance amplifier and that the DC component cannot flow away across capacitor  56 . Accordingly, while the capacitor  56  deviates alternating current to the guard node  22 , it does not reduce the sensitivity of the measurement electronics regarding the unknown impedance  16 . Resistor  60  substantially defines the gain (or transimpedance) of the transimpedance amplifier  50 . Resistor  62  and capacitor  64  ensure that the open-loop gain of the amplifier  66  and its feedback components has enough phase margin, preventing oscillation of the feedback system. 
     The low-pass filter with gain and offset correction  52  has as inputs the output of the transimpedance amplifier  50  and a third PWM output  68  of microcontroller  30 . The output voltage of transimpedance amplifier  50  is typically too small to be directly read by the ADC (analog-to-digital converter) input  70  of microcontroller  30 . Also, there is still an appreciably large amount of the guard sine wave present at the output  67  of the transimpedance amplifier  50 . Therefore, the low pass filter with gain and offset correction  52  removes the residual guard sine wave at the output  67  of transimpedance amplifier  50  by low-pass filtering and amplifies the resulting DC voltage with a fixed gain. In order to avoid that the amplification leads to a DC voltage overdriving the ADC input  70  of microcontroller  30 , an offset correction is applied by injecting a signal from the PWM output  68  of the microcontroller  30  into the input of low-pass filter with gain and offset correction  52 . The resistance ratio between resistors  72  and  74  defines the offset correction range. Resistors  72 ,  74 ,  76  and  78 , capacitors  80 ,  82  and the operational amplifier  84  form together a so-called multiple feedback (MFB) second order active low-pass filter. The DC voltage source  86  defines the DC operating point of the low pass filter with gain and offset  52 . Typically, the DC voltage of DC voltage source  86  is chosen equal to one half of the supply voltage of the microcontroller  30 . 
     For calibration purposes, a reference impedance (also: normative impedance)  88  is arranged, in parallel to the unknown impedance  16 , between the sense node  44  and circuit ground. The reference impedance is arranged in series with a switch  90 , which is controlled via a further digital output  92  of the microcontroller  30 . The microcontroller  30  periodically opens and closes the switch  90  (e.g. at a rate between 10 Hz and 10 kHz) and thereby switches reference impedance  88  in parallel to the unknown impedance  16 . For each switch position of switch  90 , the microcontroller  30  measures a complex value. Let α denote the complex value obtained when the switch  90  is open and β the complex value obtained when the switch  90  is closed. α is obtained by calculating α=α 1 +jα 2 , where α 1  is the voltage measured at microcontroller input  70  when the phase of the multiplexer  48  with respect to the guard sine wave is φ (φ need not be known, but it must be constant) and α 2  is the voltage measured at microcontroller input  70  when the phase of the multiplexer  48  with respect to the guard sine wave is φ+90°. For the measurements of both α 1  and α 2  the switch  90  is open. β is obtained by calculating β=β 1 +jβ 2 , where β 1  is the voltage measured at microcontroller input  70  when the phase of the multiplexer  48  with respect to the guard sine wave is φ (same value as for the determination of α) and β 2  is the voltage measured at microcontroller input  70  when the phase of the multiplexer  48  with respect to the guard sine wave is φ+90°. For the measurements of both β 1  and β 2  the switch  90  is closed. The difference β−α between these two measured complex values corresponds to the complex impedance of the reference impedance  88 , because the reference impedance  88  is in parallel with the unknown impedance  16 , which will be denoted Z X  in the following. Since the complex impedance, Z REF , of reference impedance  88  is known, the ratio between the difference β−α and the known complex value Z REF  of the reference impedance can be used as a calibration factor. The microcontroller  30  thus divides the measured complex value α by the difference value β−α and multiplies the result with the known value Z REF  of the reference impedance  88 . (This can be deduced as follows: Z X /Z REF =α/(β−α), therefore, Z X =Z REF α/(β−α).) 
     In the illustrated embodiment, a second reference impedance  94  is provided. A multiplexer  96  connects it alternately to the current input  54  of the transimpedance amplifier or to the guard node  22 . On the other side, the second reference impedance  94  is connected to circuit ground via a switch  98 . The switching action of the multiplexer  96  is controlled by a fourth PWM output  100  of the microcontroller  30 . 
     When the above-described measurements of α and β are carried out, the microcontroller  30  preferably keeps the second reference impedance  94  connected to the guard node  22 , so as to avoid that it enters as an offset into the measurement. 
     The second reference impedance  94  is preferably used in the following way. First, the multiplexer  48  is controlled such that it switches the sense electrode  12  to the guard node  22 . The measurement electronics is thereby rendered insensitive to any changes of the unknown impedance  16 . Via a further digital output  102 , the microcontroller  30  closes the switch  98 . The multiplexer  96  is then controlled to alternatingly connect the second reference impedance  94  to the guard node  22  during one half of the period of the guard voltage and to the current input  54  of the transimpedance amplifier  50  during the other half of the period of the guard voltage. The switching action of the multiplexer  94  is controlled by the fourth PWM output  100  of the microcontroller  30 . The frequencies of the signals on the PWM outputs  28  and  100  are identical during this measurement and the software of the microcontroller  30  controls the phase shift between these signals. The microcontroller  30  alternates the phase shift between two phase shift values that are preferably 90 degrees apart. Measuring the current flowing into the second reference impedance  94  for each of these phase shift values yields two real-valued measures, herein denoted γ 1  and γ 2 , or one complex value γ=γ 1 +jγ 2 , which corresponds to the complex impedance of the second reference impedance. This arrangement has the advantage that the reference impedance  94  is measured individually and independently from the unknown impedance. Assuming that the values α and γ have been obtained using the same phase offset φ, the microcontroller can calculate the unknown capacitance can as Z X =Z REF2 α/γ, where Z REF2  is the known value of the second reference impedance  94 . It is worthwhile noting that the second reference impedance  94  can be used as an alternative or a complement to the first reference impedance  88 . Implementing both (or even further) reference impedances adds redundancy to the system and may be useful if a high fault detection probability is required. 
     Additionally, the sum of the offset capacitance between the common node and ground and the offset capacitance between output and ground of the multiplexer  96  can be measured, assuming that the parasitic capacitance of the multiplexer  96  is significantly less than the offset capacitance. For this measurement, the microcontroller  30  opens switch  98  but otherwise keeps the same system configuration as for the measurement of γ. The measurement yields in this case a complex value, hereinafter denoted γ′, which corresponds to the offset impedance (or the offset capacitance) of the multiplexer  96 . Assuming that all the multiplexers used in the measurement electronics are substantially identical, the parasitic offset impedances thereof can be subtracted from the measured unknown impedance for a higher accuracy of the measurement. 
     As also illustrated in  FIG. 1 , the capacitive sensing device  10  may comprise a plurality of sense electrodes and/or guard electrodes. A second sense electrode  12   a  is connected to a second sense node  44   a  via the core conductor  46   a  of a shielded cable  20   a . A multiplexer  48   a  is provided for connecting the sense node  44   a  alternatingly to the current input  54  of the transimpedance amplifier  50  and to the guard node  22 . When the capacitive sensing device  10  measures the first unknown impedance  16  individually, the microcontroller  30  controls the multiplexer  48   a  such that the second sense electrode  12   a  is connected to the guard node  22 . The microcontroller  30  controls the multiplexer  48   a  with a fifth PWM output  28   a . The sensitivity of the second sense electrode  12   a  is reduced towards one side by means of a second shield electrode  14   a , which is connected to the guard node  22  via the shield conductor  24   a  of the shielded cable  20   a . EMI protection capacitor  58   a , together with capacitor  38 , guarantees that high-frequency currents may flow to circuit ground substantially without perturbing the transimpedance amplifier  50  and the other measurement electronics. When the capacitive sensing device  10  measures the second unknown impedance  16   a  individually, the microcontroller  30  controls the multiplexer  48  such that the first sense electrode  12  is connected to the guard node  22  while it operates the multiplexer  48   a  as it has been described above for the first multiplexer  48 . 
     Whereas only to two antenna arrangements are shown in  FIG. 1 , it should be noted that more antenna arrangements could be connected in a similar way as shown. 
     Capacitors  38 ,  58  and  58   a  ensure that an RF current injected into the sensor cabling during the so-called BCI (bulk current injection) test substantially flows through capacitors  58  and  38  or  58   a  and  38  into circuit ground rather than into the measurement electronics. Additionally, residual RF current flowing into node  54  via the multiplexer  48  or  48   a  flows, to a large extent, into circuit ground across capacitors  56  and  38 . 
     In a preferred implementation of the embodiment illustrated in  FIG. 1 , the following system parameters have been chosen:
         resistance  32 =330Ω   capacitance  34 =2.7 nF   inductance  36 =560 μH   capacitance  38 =4.7 nF   resistance  40 =3.3 kΩ   capacitance  42 =10 nF   capacitance  56 =47 nF   capacitance  58 =470 pF   resistance  50 =100 kΩ   resistance  62 =100Ω   capacitance  64 =1 nF   resistance  72 =3.6 kΩ   resistance  74 =33 kΩ   capacitance  80 =22 nF   resistance  76 =33 kΩ   resistance  78 =10 kΩ   capacitance  82 =680 pF   impedance  88 =100 kΩ   impedance  94 =100 kΩ   capacitance  58   a= 470 pF   voltage of DC voltage source  86 =2.5 V   microcontroller supply voltage=5 V   operating frequency (i.e. frequency of the guard voltage)=125 kHz       

     While specific embodiments have been described in detail, those skilled in the art will appreciate that various modifications and alternatives to those details could be developed in light of the overall teachings of the disclosure. Any numerical values indicated herein are also provided only for the purpose of illustration. Accordingly, the particular arrangements disclosed are meant to be illustrative only and not limiting as to the scope of the invention, which is to be given the full breadth of the appended claims and any and all equivalents thereof.