Abstract:
A channel matrix is estimated for a channel subject to frequency and time selective fading in a multiple-input multiple-output (MIMO) wireless communication network. The method uses a recursive least-squares (RLS) process and an expectation-maximization (EM) process, based on a L 1 -L q  regularized sparse regression. The method significantly improves an accuracy of the channel estimation by using high-order matrix extension to obtain covariance matrices. The covariance matrices and an initial estimate are refined by a sparse EM to obtain the final estimate of the channel.

Description:
FIELD OF THE INVENTION 
       [0001]    This invention relates generally to communication networks, and more particularly to estimating channels between transceivers in wireless multiple-input multiple-output (MIMO) networks. 
       BACKGROUND OF THE INVENTION 
       [0002]    Wireless communication experiences non-stationary, fast fading channel conditions, especially for transceivers that move rapidly. This makes it difficult to estimate the channel to enable coherent signal detections. 
         [0003]    Many channel estimation methods for frequency and time selective fading are known. One of the most widely used methods performs a recursive least-square (RLS) process, which has a good tracking ability and low complexity. The tracking performance can be improved with an order extended RLS process. 
         [0004]    While the order extended RLS process offers good estimation accuracy in a high signal-to-noise ratio (SNR) regimes and for very fast fading channels, the estimation capability is severely degraded when the channel is slow fading, sparse in frequency, and very noisy. 
         [0005]    For sparse channel estimations, a sparse RLS process, which uses compressive sensing based on L 1 -norm regularization, is known. Compressive sensing determines sparse solutions for under-determined linear systems. However, that method assumes a single-antenna at the transceiver. In addition, the performance degrades in a very rapid fading channel because that method is based on conventional zero-order channel estimation. 
       SUMMARY OF THE INVENTION 
       [0006]    The embodiments of the invention provide a method for estimating a channel in a frequency and time selective multiple-input multiple-output (MIMO) network, wherein transceivers have multiple antennas. The method estimates a rapid time-varying channel with channel sparseness in the time domain. 
         [0007]    The invention solves the following three problems that are characteristic of prior art channel estimation methods. The estimation accuracy is degraded for low SNR regimes when high-order estimations are used. Low-order estimation suffers from a severe performance degradation for high SNR regimes in time-varying channels. Sparse channel estimations are computationally complex. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0008]      FIG. 1  is a block diagram of a multiple-input multiple-output (MIMO) wireless communication network according to embodiments of the invention; 
           [0009]      FIG. 2  is a time diagram of wireless transmission for channel estimation according to embodiments of the invention; 
           [0010]      FIG. 3  is a schematic of a frequency-and-time selective fading MIMO channel with order-extended polynomial expression according to embodiments of the invention; 
           [0011]      FIG. 4  is a block diagram of the high-order sparse MIMO channel estimation method employing an order-extended recursive least-squares (RLS) procedure, and an expectation-maximization (EM) procedure according to embodiments of the invention; 
           [0012]      FIG. 5  is a block diagram of the order-extended RLS procedure for estimating the time-varying channel according to embodiments of the invention; and 
           [0013]      FIG. 6  is a block diagram of a sparse EM procedure according to embodiments of the invention. 
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       [0014]    As shown in  FIG. 1 , the embodiments of the invention provide a method  400  for estimating a channel  130  in a frequency and time selective multiple-input multiple-output (MIMO) network  100 . The network includes at least two transceivers. The first transceiver can operate as a transmitter  141 , while the transceiver operates as a receiver  142 , or vice versa. The steps of the method  400  operate in a processor at the receiver. The processor is connected to memory and input/output interfaces. 
         [0015]    The transmitter includes a set of transmit antennas  145 , and the receiver includes a set of receive antennas  146 . Let M and N be the number of transmit antennas and that of receive antennas, e.g., two each, M=N=2. An M-dimensional vector signal x  121  is transmitted by the antennas at the transmitter. The corresponding received signal  122  at the receiver, after passing through the channel  130 , is y, which is an N-dimensional vector. The channel can be represented by a matrix H. At the receiver, an estimated version of the channel matrix H′  131  is obtained by the steps of the estimation method  400 . 
         [0016]      FIG. 2  shows a detailed time diagram of the signal transmission x  121  from the transmitter to the receiver for the channel estimation. The transmitter sends a transmitting sequence x( 1 )  201  through x(k 0 )  202 , where k 0  denotes a length of the packet sequence in symbols. The receiver obtains the corresponding sequence from y( 1 )  211  to y(k 0 )  212 . The frequency-selective fading channel has a memory incurring inter-symbol interference. Let P be a maximum channel memory length. At the k th  symbol, the received signal y(k)  213  is expressed by a mixture of the previous P+1 symbols through the frequency-selective fading channel H(k)  221  as follows: 
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         [0000]    where H p (k) is the p th  delayed-tap channel matrix of size N×M, z(k) is the receiver noise of size N by 1, H(k) is the compound channel matrix of size N×M(P+1), and x′(k) is the compound transmit signal, which stacks the previous P+1 symbols. The channel rapidly changes in time  222 . Hence, the channel is termed a frequency-and-time selective channel. 
         [0017]      FIG. 3  shows the time variation of the MIMO channel, H(k)  221 , in symbol time index k. For most typical radio channels, there are only a few non-zero entries in the channel matrix H(k)  221  and many zero or near zero values, and the Doppler frequency that causes time variation is bounded by the maximum Doppler frequency. This implies that the channel H(k)  221 , which spans in space dimension  301 , in frequency (or, delay domain) dimension  302 , and in time dimension  303 , becomes very sparse. 
         [0018]    Sparsity: In the field of numerical analysis, a sparse matrix is a matrix populated primarily with many zero or near zero values. 
         [0019]    The method  400  estimates an optimal frequency-and-time selective channel matrix H(k), using the high-order sparse regressions for such a sparse channel matrix. 
         [0020]    Introducing the Taylor-series expansion, the time-varying channel matrix can be expressed by high-order polynomial  311  as follows: 
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         [0000]    where G d (k)  312  is the d″ order polynomial term of size N×M(P+1), D is a maximum polynomial order under consideration, G(k) is an N×M(P+1)(D+1) dimensional matrix which stacks G 0 (k), G 1 (k), and G D (k), and D′(k) is an order extension matrix of size M(P+1)(D+1)×M(P+1). This order-extended polynomial expression reduces the effective fading speed by considering a new channel matrix G(k), rather than the original channel matrix H(k) because the major time variation is absorbed by the order extension matrix D′(k). Moreover, the new channel matrix G(k) is sparser than the channel matrix H(k) because of an additional order dimension. Note that the use of another order extension, such as Fourier-series expansion, provides an alternative embodiment of the invention. 
         [0021]      FIG. 4  shows the details of the high-order sparse MIMO channel estimation method  400 , which employs the order-extended recursive least-squares (RLS) procedure  500  and the expectation-maximization (EM) procedure  600  for low-complexity process at the receiver  142 . The steps of the method include six major blocks: initialization  401 , order extension  402 , RLS  400 , EM  500 , order folding  403 , and loop control  404 . 
         [0022]    At the initialization step  401 , some parameters (polynomial order D, forgetting factor r, an EM constant a, and sparseness control factor g) used in the RLS procedure  500  and EM procedure  600  are set initialized, e.g., D=2, r=0.98. The values of a and g are adjusted by the channel signal-to-noise ratio (SNR), more specifically is exponentially decreasing with SNR, and g is linearly increasing with SNR. The initialization step  401  also sets up some matrices (transmit signal memory x′(0)=0, an order-extended channel G′(0)=0, a signal correlation matrix R, EM correlation matrices A=0 and B=0) required for RLS and EM procedures. The correlation matrix R is initialized as R=cI with large constant c and identity matrix I, e.g. R=100I. The symbol index is initialized as k=1. 
         [0023]    At the order extension step  402 , the receiver obtains the received signal y(k) and the transmitted signal x(k). The transmitted signal x(k) is a packet including a training sequence and a data sequence. The training sequence is known at the receiver during a channel training period (typically in the first several symbols (preamble) of the packet), while x(k) is given by a decision through decoding process at the receiver during the data transmission period. The transmitted signal x(k) is buffered in a M(P+1)−dimensional memory as x′(k). The buffered signal x′(k) is extended to D th  order polynomial signal u=D′(k)×′(k) of size M(P+1)(D+1). 
         [0024]    In the RLS procedure  500 , the least-squares solution of the high-order channel matrix G′(k), is recursively obtained given y(k), u, R, and r. The RLS procedure is used to accelerate the convergence speed of the following EM procedure. It is possible to exclude the RLS step by duplicating as G′(k)=G′(k−1). In the EM step procedure, the estimate G′(k) is iteratively refined to take advantage of channel sparseness through the expectation and maximization procedures, given y(k), u, r, a, g, A, and B. 
         [0025]    After the convergence in the EM procedure, the order folding step  403  obtains the optimal channel estimation matrix H′(k) from the order-extended channel G′(k), as G′(k)D′(k). In the loop control step  404 , the symbol index k is incremented and the steps from the order extension  402  are iterated until k reaches the end of the sequence k 0    405 . 
         [0026]      FIG. 5  shows the high-order RLS procedure  500 . Given initialized parameters (r and R)  401  and order-extended signal u  402 , the least-squares estimate is recursively obtained by the following four steps; determining a forward error vector  501 , determining an update vector  502 , updating a correlation matrix R  503 , and updating the order-extended channel matrix  504 . 
         [0027]    First, the forward error vector e of size N is determined by subtracting an expected received replica G′(k−1) u from the received signal y(k) as follows: 
         [0000]        e=y ( k )− G ′( k− 1) u,  
 
         [0000]    where the error vector e is used as a gradient descent to minimize the mean-square error. Next, the updating vector q of size M(P+1)(D+1) is determined as follows: 
         [0000]    
       
      
       q=Ru/r,  
      
     
         [0000]    which is used as a step size of the gradient descent. The correlation matrix R is then updated as 
         [0000]        R←R/r−qq* /(1+ q*u ), 
         [0000]    where the super-script [.]* denotes the Hermitian transpose of a matrix. This updated correlation matrix is used at the next symbol to generate a new updating vector q. Finally, the order-extended channel matrix G′(k) is obtained from the previous value G′(k−1) and the gradient vectors e and q as follows: 
         [0000]        G′ ( k )= G′ ( k− 1)+ eq*.    
         [0028]    This high-order RLS method obtains the optimal estimate, which minimizes the mean-square error, using high-order polynomial regressions. However, it does not consider the sparseness underlying in the high-order channel matrix. The channel matrix is further refined by the sparse EM procedure  600 . 
         [0029]      FIG. 6  shows the steps of the sparse EM procedure. At the first step  601 , an auto-correlation matrix A and a cross-correlation matrix B are updated, given u and y(k) with pre-determined constants a and r, as follows: 
         [0000]    
       
      
       A←rA+auu*  
      
     
         [0000]        B←rB+ay ( k ) u*.    
         [0030]    At the second step  602 , the high-order channel matrix G(k) is expected to obtain an expectation value S by considering the matrix sparseness as follows 
         [0000]        S=G′ ( k )( I−A )+ B.    
         [0031]    Given the expectation S, the high-order channel matrix G(k) is refined to maximize a likelihood for sparse matrices at the maximization step  603 , as follows: 
         [0000]        G′ ( k )← F   th ( S,ag ),
 
         [0000]    where g is a sparseness control factor, which is initialized at the step  401 , and F th (•) is a soft-threshold function defined as 
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         [0000]    with |•| being the magnitude of a complex number. If the high-order channel matrix G′(k) is not sufficiently converged  604 , the expectation step  602  and the maximization step  603  are iterated  605  until the convergence. Otherwise, the EM step ends, and the optimal sparse channel estimation G′(k) is obtained. 
       EFFECT OF THE INVENTION 
       [0032]    Compared to the conventional channel estimation methods, the invention has the following advantages. 
         [0033]    The method does not require pre-estimation to determine non-zero tap positions. 
         [0034]    The method automatically controls the optimal order, the optimal tap gains, and the optimal tap for time-varying MIMO channels. 
         [0035]    The method provides near-optimal performance for any channel conditions over the whole SNR regimes, any dense-or-sparse channels, and channel changing speed, whereas the conventional method has a poor performance particularly for high SNR regimes and dense channels. 
         [0036]    The method can be applied to other networks including wired communications, fiber-optics communications, and control networks which can be modeled by frequency-and-time selective MIMO tapped delay line. Here, MIMO includes any single-antenna scenarios as well as multiple-antenna cases. 
         [0037]    Although the invention has been described with reference to certain preferred embodiments, it is to be understood that various other adaptations and modifications can be made within the spirit and scope of the invention. Therefore, it is the object of the append claims to cover all such variations and modifications as come within the true spirit and scope of the invention.