Abstract:
A digital subscriber line (DSL) driver allows a transmitter to monitor its own transmit spectrum at the subscriber loop and adjust the transmit spectrum based on detected line conditions, affected by the presence of bridged taps or any other impedance variations. The transmit spectrum is preferably equalized so that all carriers, or tones, transmit using the same power and exhibit the same margin. The invention is applicable to DMT and single carrier modulation formats.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
   This application is a continuation of application Ser. No. 09/733,841, filed Dec. 7, 2000 now U.S. Pat. No. 6,975,694, which claims the benefit of U.S. Provisional Application No. 60/170,988, filed Dec. 15, 1999. These applications are entirely incorporated herein by reference. 

   TECHNICAL FIELD 
   The present invention relates generally to communications systems, and more particularly, to a subscriber line driver for a digital subscriber line (DSL) communications system. 
   BACKGROUND OF THE INVENTION 
   Data communication typically occurs as the transfer of information from one communication device to another. This is typically accomplished by the use of a modem located at each communication endpoint. In the past, the term modem denoted a piece of communication apparatus that performed a modulation and demodulation function, hence the term “modem”. Today, the term modem is typically used to denote any piece of communication apparatus that enables the transfer of data and voice information from one location to another. For example, modern communication systems use many different technologies to perform the transfer of information from one location to another. Digital subscriber line (DSL) technology is one vehicle for such transfer of information. DSL technology uses the widely available subscriber loop, the copper wire pair that extends from a telephone company central office to a residential location, over which communication services, including the exchange of voice and data, may be provisioned. The subscriber loop includes two wires, which are commonly referred to as “tip” and “ring.” Unfortunately, the subscriber loop may also include “bridged taps,” which are unterminated subscriber loops and which present high impedance to communication devices coupled to the line. 
   DSL devices can be referred to as modems, or, more accurately, transceivers, which connect the telephone company central office to the user, or remote location, typically referred to as the customer premises (CP). DSL communication devices use different formats and different types of modulation schemes and achieve widely varying communication rates. However, even the slowest DSL communications devices achieve data rates far in excess of conventional point-to-point modems. 
   Some of the available modulation schemes include quadrature-amplitude modulation (QAM), carrierless amplitude/phase (CAP) and discrete multi-tone (DMT) modulation. In a DMT modulation scheme, a number of carriers, commonly referred to as “tones” are encoded with the information to be transmitted and communicated over the communications channel. This information in the form of data words is encoded into signal space constellations and then transmitted. In a typical DMT transmitter, 256 carrier tones are used to encode the data and are added together resulting in a very high peak signal power due to the high numerical peak resulting from the addition of the 256 tones. Contributing to this peak power is the DMT algorithm, which allows the power on individual tones, or carriers, to be increased by up to 2.5 dB to satisfy margin requirements. The number of bits encoded into a symbol on each carrier is selected to bring margin to within 3 dB of the specified margin. The margin is then further improved on selected carriers by boosting their transmit signal power. Unfortunately, this boost in power increases the peak transmit signal power. 
   Power consumption is further increased due to the use of square signal space constellations used in conventional DMT transmitters and the allowed +2.5 dB to −14.5 dB power variation allowed on each carrier tone. Square signal space constellations have an inherently high peak signal power due to the location of the highest power signal point. The peak signal power in conventional DMT transmitters is sufficiently high to cause saturation or clipping of the transmitter in normal operation, thus preventing the use of DMT based systems in dense central site locations. Conventional DMT allows a probability of clipping of 10 −7 . To combat this inherent deficiency, conventional DMT transmitters use expensive Reed-Solomon forward error correction encoders combined with bit-wise interleavers. Coders such as these induce a significant amount of throughput delay and are unsuitable for multi-point communication environments. 
   Another limitation of DMT is a lack of transmit equalization or pre-emphasis. An assumption in DSL is that a flat transmit spectrum is optimum given that cross-talk from similar systems is the primary line impairment. Unfortunately, this assumption disregards the unique impedance of each subscriber loop and may result in a non-flat transmit spectrum on the tip and ring wires of the subscriber loop. 
   The number of bits encoded on each DMT carrier is selected in whole bit increments to bring the margin associated with each tone to within 3 dB of the specified margin. Contributing to the peak power problem, the DMT modulation scheme allows the power on individual tones to be increased by up to 2.5 dB to satisfy margin requirements. While increasing the power on some tones, the system reduces the power on other tones to maintain the specified transmit power. This scheme of tone power variation is useful in instances where it may be desirable to turn off specified tones and allocate their power to other tones. Unfortunately, this tone powers variation results in spectrum management difficulties. Conventional DMT systems simply turn off specified tone carriers and increase others by the allowed 2.5 dB. Unfortunately, this results in the undesirable situation in which some carriers will be 2.5 dB hotter than necessary in certain spectral bands, resulting in undesirable cross-talk, while other carrier tones are switched off completely. The tones at the high end of the frequency spectrum are frequently switched off. 
   Cross talk between wire pairs in cable bundles or in dense equipment cabinets is the major contributor to degradation in DSL networks. Spectrum management schemes have been proposed to limit cross-talk by specifying a Power Spectrum Density (PSD) for all DSL transmitters. DMT allows PSD pass band ripple of +3.5 dB to accommodate the +2.5 dB power variation on individual tones. While currently allowed, a boost of 2.5 dB in power will effectively lower channel capacity by 2.5 dB on neighboring wire pairs, thereby nullifying any true gain that may have been anticipated. It is desirable to have a DMT system that can transmit a specified PSD with pass band ripple of less than 1 dB. 
   Thus, it would be desirable to have a DMT communication system capable of providing transmit spectrum equalization, thus reducing or eliminating the need to boost power on individual carriers. 
   SUMMARY 
   The invention is an improved system and method for communicating over a digital subscriber line. 
   The invention allows a transmitter to monitor its own transmit spectrum at the subscriber loop and adjust the transmit spectrum based on detected line conditions. The transmit spectrum is preferably equalized so that all carriers, or tones, transmit using the same power and exhibit the same margin. The invention is applicable to DMT and single carrier modulation formats. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The invention can be better understood with reference to the following drawings. The components in the drawings are not necessarily to scale, emphasis instead being placed upon clearly illustrating the principles of the present invention. Moreover, in the drawings, like reference numerals designate corresponding parts throughout the several views. 
       FIG. 1A  is a block diagram illustrating the digital subscriber line communication environment in which the present invention resides; 
       FIG. 1B  is a graphical representation illustrating the frequency spectrum of the communication that occurs between the central office and the customer premises of  FIG. 1A ; 
       FIG. 2  is a block diagram illustrating the DSL transceiver of  FIG. 1A ; 
       FIG. 3  is a block diagram illustrating the transmitter of  FIG. 2 , including the power and spectrum control feedback loop of the invention; 
       FIG. 4  is a block diagram illustrating an alternative embodiment of the transmitter of  FIG. 2 ; 
       FIG. 5  is a block diagram illustrating the receiver  400  of  FIG. 2 ; 
       FIG. 6A  is a graph illustrating a conventional DMT transmit spectrum; and 
       FIG. 6B  is a graph illustrating a DMT transmit spectrum after operation of the power and spectrum control feedback loop of  FIG. 3 . 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   Although described with particular reference to an asymmetric digital subscriber line (ADSL) communication system using DMT modulation, the digital subscriber line driver of the invention can be implemented in any communication system. 
   Furthermore, the digital subscriber line driver can be implemented in software, hardware, or a combination thereof. In a preferred embodiment(s), selected portions of the digital subscriber line driver are implemented in hardware and software. The hardware portion of the invention can be implemented using specialized hardware logic. The software portion can be stored in a memory and be executed by a suitable instruction execution system (microprocessor). The hardware implementation of the digital subscriber line driver can include any or a combination of the following technologies, which are all well known in the art: a discrete logic circuit(s) having logic gates for implementing logic functions upon data signals, an application specific integrated circuit (ASIC) having appropriate logic gates, a programmable gate array(s) (PGA), a field programmable gate array (FPGA), etc. 
   Furthermore, the digital subscriber line driver software, which comprises an ordered listing of executable instructions for implementing logical functions, can be embodied in any computer-readable medium for use by or in connection with an instruction execution system, apparatus, or device, such as a computer-based system, processor-containing system, or other system that can fetch the instructions from the instruction execution system, apparatus, or device and execute the instructions. 
   In the context of this document, a “computer-readable medium” can be any means that can contain, store, communicate, propagate, or transport the program for use by or in connection with the instruction execution system, apparatus, or device. The computer readable medium can be, for example but not limited to, an electronic, magnetic, optical, electromagnetic, infrared, or semiconductor system, apparatus, device, or propagation medium. More specific examples (a nonexhaustive list) of the computer-readable medium would include the following: an electrical connection (electronic) having one or more wires, a portable computer diskette (magnetic), a random access memory (RAM), a read-only memory (ROM), an erasable programmable read-only memory (EPROM or Flash memory) (magnetic or electronic), an optical fiber (optical), and a portable compact disc read-only memory (CDROM) (optical). Note that the computer-readable medium could even be paper or another suitable medium upon which the program is printed, as the program can be electronically captured, via for instance optical scanning of the paper or other medium, then compiled, interpreted or otherwise processed in a suitable manner if necessary, and then stored in a computer memory. 
   Turning now to the drawings,  FIG. 1A  is a block diagram illustrating the digital subscriber line communication environment  100  in which the present invention resides. The DSL communication environment  100 , for illustrative purposes only, is an asymmetric digital subscriber line (ADSL) communication environment. However, other DSL communication environments are contemplated by the invention. For example, the invention contemplates the use of symmetric DSL technologies such as high bit rate DSL (HDSL), symmetric DSL (SDSL), and multi-rate SDSL (MSDSL). Furthermore, the invention contemplates other asymmetric digital subscriber line technology such as rate adaptive DSL (RADSL). 
   Central office  102  connects to customer premises  108  via communication channel  106 . Communication channel  106  can be the copper wire pair that typically extends between central office  102  and a remote customer location, and is commonly referred to as the “local loop,” or “subscriber loop.” For exemplar purposes only, the transmission scheme between central office  102  and customer premises  108  is point-to-point full duplex ADSL, using discreet multi-tone (DMT) line coding. Other communication schemes are possible between central office  102  and customer premises  108 , such as variations of multi-point and half duplex. 
   Central office  102  includes a number of DSL transceivers, an exemplar of one of which is illustrated using reference numeral  200 . DSL transceiver  200  exchanges information with data terminal equipment (DTE)  101  via connection  104  and interfaces to communication channel  106  in order to communicate with DSL transceiver  114  located at customer premises  108 . DTE  101  can be, for example, a computer to which the DTE  110  in customer premises  108  is communicating, or can represent access to the Internet. 
   DSL transceiver  114  located at customer premises  108  connects to DTE  110  and plain old telephone service (POTS) device  112  connects directly to communication channel  106 . Although omitted for clarity, a POTS splitter is typically installed between the communication channel  106  and POTS device  112  to prevent the POTS device  112  from interfering with the DSL communication and to prevent DSL signals from degrading POTS performance. DTE  110  can be a personal computer and POTS device  112  can be a typical telephone. Typically, the DSL transceiver  114  is located in a home or office, and is used to allow simultaneous data communication (using DTE  110 ) and voice communication (using POTS device  112 ). 
   DSL transceiver  114  communicates over communication channel  106  with DSL transceiver  200  in order to exchange data information. Simultaneously with the data, voice information is transmitted over communication channel  106  and separated by DSL transceiver  200  via connection  107  and sent to typical telephone company POTS switching equipment (not shown). Data is exchanged between customer premises  108  and central office  102  in order to allow DTE  110  to access, for example, DTE  101 , which may be an Internet access device provided by an internet service provider (ISP). 
   Although illustrated using a single customer premises  108  connected to central office  102 , typically a number of customer premises locations will be connected to one central office  102  using a plurality of communication channels. Furthermore, it is also possible for a number of customer premises locations to be connected to one DSL transceiver located at a central office. However, for purposes of explanation, the invention will be described with reference to a single customer premises  108  communicating with a single DSL transceiver  200  located at central office  102 . Furthermore, the invention described hereafter is embodied in both DSL transceiver  200  located at central office  102  and in DSL transceiver  114  located at customer premises  108 . However, for ease of illustration, only a single DSL transceiver  200  will be described below. 
     FIG. 1B  is a graphical representation illustrating the frequency spectrum of the communication that occurs between central office  102  and customer premises  108  of  FIG. 1A . The vertical axis of graph  120  represents transmission power and the horizontal axis of graph  120  represents frequency. Generally, the communication that occurs between DSL transceiver  200  and DSL transceiver  114  is divided by frequency, with the transmission from central office  102  toward customer premises  108  (downstream) occurring at a frequency higher than the transmission of information from DSL transceiver  114  located at customer premises  108  toward the DSL transceiver  200  located at central office  102  (upstream). For example, the curve indicated using reference numeral  122  represents upstream transmission from the customer premises  108  toward the central office  102 , and the curve indicated using reference numeral  126  represents downstream transmission occurring from the central office  102  toward the customer premises  108 . Transmission from the customer premises  108  toward the central office  102  is generally lower in frequency in order to avoid cross-talk that may occur as the signal gets nearer to the central office  102  and an increasing number of communication channels, such as channel  106 , are bundled together in common cable bundles. Furthermore, although shown as a discreet difference in frequencies indicated by region  124  in  FIG. 1B , it is possible that there is some overlap occurring between the curves  122  and  126 . Further still, although shown as symmetric in nature for simplicity, the curves  122  and  126  may not appear identical. For example, the curve  122  might be higher in power and occur over a shorter frequency spectrum than curve  126 . 
     FIG. 2  is a block diagram illustrating the DSL transceiver  200  of  FIG. 1A . The DSL transceiver  200  communicates with DTE  101  via connection  104  where data supplied to and received from DTE  101  is exchanged via input/output element  202  within DSL transceiver  200 . DSL transceiver  200  also includes microprocessor  201  and memory  204  in communication via bus  207  with digital signal processor (DSP)  208 . The memory  204  and the microprocessor  201  work in cooperation to store and execute the logic of the invention. In the absence of a microprocessor, the logic of the invention may be executed in the DSP  208 . Although DSP  208  as illustrated includes the transmitter  300  and receiver  400  of the invention, the transmitter  300  and receiver  400  may be implemented separately. The receiver  400  includes the power and spectrum control feedback loop  450 , which detects the transmit symbol at the output of the transmitter. The power and spectrum control feedback loop supplies carrier tone scaling signals via connection  338  to the transmitter  300  and will be described in greater detail below. Although shown for simplicity in  FIG. 2  as a single connection, the connection  338  can be a plurality of connections, corresponding to the plurality of carrier tones modulated by the transmitter  300 . The power and spectrum control feedback loop  450  can be considered part of a line driver, and resides in the receiver, but is used by the transmitter  300  during an initial transmission. The power and spectrum control feedback loop  450  is used to measure the power and analyze the transmit spectrum so that each of the carrier tones may be scaled, or optimized, to provide an efficient transmit spectrum. 
   Transmitter  300  communicates via connection  209  with line interface  212  in order to gain access to communication channel  106 . Similarly, receiver  400  receives signals from line interface  212  via connection  211 . 
     FIG. 3  is a block diagram illustrating the transmitter  300  of  FIG. 2 . A data signal originating in DTE  101  is communicated via bus  207  to scrambler  301 . The signal on connection  207  is in the form of a serial bit stream and the scrambler  301  provides a scrambled serial bit stream on connection  302 . The scrambler  301  can be either a self-synchronized scrambler or a preset free running scrambler as is known in the art. Depending upon the application, the preset scrambler may have some advantages, as in the case where Reed-Solomon coding is used. The scrambler  301  provides a scrambled m bit word on connection  302  to Reed-Solomon (RS) encoder  304 . The RS encoder  304  operates on the scrambled m bit word on connection  302  to provide forward error correction and provides the Reed-Solomon encoded serial bit stream including the RS encoded m bit words on connection  306 . The RS encoded serial bit stream on connection  306  is operated on by RS interleaver  307  to provide an interleaved Reed-Solomon encoded serial bit stream on connection  308 . 
   The serial bit stream on connection  308  is then supplied to tone-ordering element  309 . Tone-ordering element  309  orders the groups of bits contained in the serial bit stream on connection  308  into words that are to be encoded for each signal space constellation. Essentially, the tone-ordering element  309  groups the serial bit stream into bit-wise words, which will be encoded on each tone of the DMT transmission system. The output of the tone-ordering element  309  could be serial or parallel. In this embodiment, serial words are preferred. The tone-ordering element  309  precedes the fractional encoder  312 , however, the order of these elements may be reversed. 
   The serial words on connection  311  are then sent to fractional encoder  312 . Fractional encoder  312  is optional and can be used to encode a fractional, or non-integer number of bits onto each tone of the DMT transmitter  300  in accordance with that disclosed in commonly assigned, co-pending U.S. patent application Ser. No. 09/717,436, entitled “FRACTIONAL BIT RATE ENCODING IN A DISCRETE MULTI-TONE COMMUNICATION SYSTEM”. Fractional encoder  312 , if included, includes the logic necessary to encode fractional bits onto each of the n tones in the DMT communication scheme. The fractional encoder  312  operates on groups of words to encode a fractional (i.e., non-integer) number of bits on each tone. The fractional encoder  312  generates n constellations for n carrier tones simultaneously, in order to add a fraction of a bit to each carrier tone. 
   If the fractional encoder  312  is used, the fractionally encoded bit stream is supplied via connection  314  to constellation encoder  316 . If the fractional encoder  312  is omitted, the output of the tone ordering element  309  is sent to the constellation encoder  316 . The constellation encoder  316  encodes each word into a constellation for one carrier tone. Preferably, each word is mapped to a multi-dimensional circular signal space constellation. Circular signal space constellations are preferable because they allow an additional 0.2 dB of shaping gain, and an approximate 1.7 dB lower peak factor. Conventional DMT transmitters, by using square signal constellations, give up 0.2 dB worth of shaping gain available through the use of circular signal space constellations. This is so due to the lower peak factor available through the use of circular signal space constellations. The advantage in channel capacity of fractional bit encoding is 0.43 dB, deriving 0.2 dB from circular constellations and 0.23 dB from modulus conversion. 
   A trellis encoder  324  communicates via connection  322  with constellation encoder  316 . The trellis encoder  324  operates sequentially on each word supplied via connection  314 . The trellis encoder encodes from one tone to the next tone. Alternatively, a single carrier trellis encoder would encode from one symbol in time to the next symbol. Because DMT modulation is performed using discrete multiple carrier tones the trellis encoder encodes from one tone to the next tone. 
   The constellation encoder  316  supplies n constellations on n carrier tones represented using reference numerals  317 - 1  through  317 - n . Although shown using only four tones in  FIG. 3 , typically constellation encoder  316  would encode 256 individual carrier tones with the 256 encoded tones comprising an encoded symbol. The constellations represent one phase and amplitude for each tone in parallel. The constellation encoder  316  encodes each word into a signal space constellation to be transmitted by the DMT transmitter. The plurality of tones, each with its respective signal space constellation, comprises a symbol that is transmitted. Each of the individual carrier tones on connections  317 - 1  through  317 -n is supplied to a respective gain stage  318 - 1  through  318 -n. The gain stages  318  (g 1 –g n ) are part of a standard DMT transmitter and are augmented by respective subscriber line driver (SLD) gain scalars  319 - 1  through  319 -n (G 1 –G n ), which will be described below. 
   The output of each SLD gain scalar  319 - 1  through  319 -n is supplied to inverse discrete Fourier transform (IDFT) element  321 . The IDFT element  321  transforms the set of all tones (in parallel or taken together at one time) into a tone sequence of samples that are serially output on connection  323 . The output on connection  323  is supplied to digital-to-analog converter (DAC)  325 . The DAC  325  converts the digital domain signals on connection  323  to the analog domain on connection  326 . The analog signal on connection  326  is amplified by differential transmit amplifier  327 , which is the line driver for the transmitter  300 , for transmission via connection  209  to the line interface  212  of  FIG. 2  for transmission via communication channel  106 . DAC  325  may include a filter (not shown) to help control the power spectrum density (PSD) for spectrum management. 
   The output of differential transmit amplifier  327  on connections  327   a  and  327   b  (commonly referred to as the “tip” and “ring” signals) is supplied to receive differential amplifier  401 . The receive differential amplifier  401  is part of the power and spectrum control feedback loop  450 , which is part of the receiver  400 . The power and spectrum control feedback loop  450  also includes an analog-to-digital converter  404 , discrete Fourier transform (DFT) element  408  and gain calculation element  410 . Although the analog-to-digital converter  404 , DFT element  408  and gain calculation element  410  within the power and spectrum control feedback loop  450  are contained within the receiver  400 , these elements are used temporarily during start-up of the transmitter  300  and before they are used in the receiver  400 . Alternatively, duplicate components can ensure that the power and spectrum control feedback loop  450  can operate at any time and indeed, may ensure that it can operate periodically throughout a communication session. 
   The receive differential amplifier  401  supplies the local amplified transmit signal (taken from the output of the transmitter  300 ) via connection  402  to the ADC  404 . The ADC  404  converts the analog receive signal to the digital domain and supplies the digital samples via connection  406  to the DFT element  408 . The DFT element  408  transforms the tone sequence of samples on connection  406  into a plurality of individual tones on connection  409 . Although illustrated using a single connection, there are a plurality of signals, each having one tone, on connection  409 . DFT element  408  may include adaptive phase and/or amplitude tracking for each tone. System timing may be recovered in DFT  408  by correlating phase changes on one pilot tone or across multiple tones. Each tone on connection  409  includes a symbol encoded in a circular signal space constellation. 
   The tones are supplied on connection  409  to gain calculation element  410 . The gain calculation element  410  is used to compute local gain scalars G 1  through G n  according to the formula G i =√{square root over (p i /t i )}, where G i  is the local gain scalar, p i  is the desired power spectral density (PSD) mask at the respective frequency and t i  is the measured power of the tone. The local gain scalar G i  ( 319 - 1  through  319 -n) follows the DMT gain scalar g i  ( 318 - 1  through  318 -n) with the resultant gain scalar being g i  G i  for each tone. In this manner, the power and spectrum control feedback loop  450  ensures that the transmit PSD at the output of the transmitter  300  matches the desired PSD independently of non-ideal loop impedance, reactance, bridged taps, or any other impedance variation in the communication line. The power and spectrum control feedback loop  450  will allow the transmitter  300  to monitor for the occurrence of non-ideal attenuation at the transmit output (i.e., the tip and ring terminals) and perform spectrum correction before transmitting. The subscriber line driver provides transmit spectrum equalization, which will improve the performance of the transmitter and reduce the occurrence of power boost on individual carrier tones. 
   During start-up, one or more DMT symbols can be transmitted prior to the specified start-up sequence. A single symbol may be adequate because the power and spectrum control feedback loop  450  will be monitoring the transmitter  300 . The resulting measured amplitude t i  of each transmitted carrier tone is used to compute the new gain scalars G i  for each tone. These new gain scalars are used throughout the remainder of the specified start-up process. The start-up process is local and need not involve communication with another DMT communication device. 
   As mentioned above, the subscriber line driver may be a one-time start-up process. However, the invention may be used to continuously monitor the quality of the transmit signal and make incremental improvements to maintain a constant PSD in response to changing line conditions, such as environmental drift of the loop properties caused by temperature, moisture, or other external stimulus that may be detectable at the local transmitter  300 . Variation that is detectable at the transmitter  300  may otherwise impact the remote receiver. The power and spectrum control feedback loop  450  also allows multi-point operation. 
   Normal DMT line coding allows a remote receiver to specify the gain stages  318 - 1  through  318 -n (g 1 –g n ) at the end of the start-up phase. After start-up, new single gain stages g i ′=g i G i  may be used during data transmission. When specified in decibels, the new gain is g i ′ dB=g i  dB+G i  dB. The gain stages G 1 –G n  remain because they were activated transparently before the start-up measurements were begun by the remote receiver. Thus, the remote receiver specifies g i  relative to G i  with the assumption that the G i  will remain active. During start-up the gain g i  is set at unity gain. 
     FIG. 4  is a block diagram illustrating an alternative embodiment  350  of the transmitter of  FIG. 2 . Transmitter  350  includes constellation encoder  351 , which is supplied with a single carrier quadrature input signal, such as a carrierless amplitude/phase (CAP) modulation or a quadrature amplitude modulation (QAM) signal, or any other single carrier modulation scheme. The constellation encoder  351  encodes the supplied bit stream into a signal space constellation on connection  352 . 
   A trellis encoder  354  communicates via connection  356  with constellation encoder  351 . The trellis encoder  354  operates sequentially on each word supplied to the constellation encoder  351  similar to that described above with respect to trellis encoder  324 . The trellis encoder encodes from one symbol in time to the next symbol. 
   The signal space constellation output of constellation encoder  351  on connection  352  is supplied to finite impulse response (FIR) filter  357 . The FIR filter  357  provides the time domain transmit symbol on a single carrier via connection  358  to the DAC  359 . The DAC  359  converts the digital signal on connection  358  to the analog domain for amplification by the transmit differential amplifier  362 . The analog signal on connection  361  is amplified by differential transmit amplifier  362  for transmission via connection  209  to the line interface  212  of  FIG. 2  for transmission via communication channel  106 . DAC  359  may include a filter (not shown) to help control the PSD for spectrum management. 
   The output of differential transmit amplifier  362  on connections  362   a  and  362   b  (commonly referred to as the “tip” and “ring” signals) is supplied to receive differential amplifier  401  as described above with respect to  FIG. 3 . However, when used with the transmitter  350  of  FIG. 4 , the power and spectrum control feedback loop  450  provides a gain calculation value on each connection  364 - 1  through  364 -n to the coefficient calculation element  366 . With single carrier modulation the gain values are calculated at any frequencies that are convenient for the DFT element  408  and span the bandwidth of the transmitter. The coefficient calculation element  366  provides the adaptive filter coefficients to the FIR filter  357  via connection  367 . Coefficient calculation methods are well known in the art. One method is a straightforward inverse discrete Fourier transform. A variation on the inverse discrete Fourier transform method is known as frequency sampling filter coefficients. Another method for calculating the coefficients is described in commonly owned U.S. Pat. No. 5,008,903 entitled “ADAPTIVE TRANSMIT PRE-EMPHASIS FOR DIGITAL MODEM COMPUTED FROM NOISE SPECTRUM,” issued on Apr. 16, 1991 to Betts et al. 
     FIG. 5  is a block diagram illustrating the receiver  400  of  FIG. 2 . A signal received on communication channel  106  is supplied via line interface  212  ( FIG. 2 ) via connection  211  to analog-to-digital converter (ADC)  404 . ADC  404  may include a filter (not shown) and/or equalizer (not shown) to help control undesirable signals and intersymbol interference. ADC  404  converts the analog signal received on connection  211  into a serial bit stream on connection  406 . The serial bit stream on connection  406  is supplied to discreet Fourier transform (DFT) element  408 . The DFT element  408  transforms the tone sequence of samples on connection  406  into a plurality of individual tones represented using reference numerals  409 - 1  through  409 -n. DFT  408  may include adaptive phase and/or amplitude tracking for each tone. System timing may be recovered in DFT  408  by correlating phase changes on one pilot tone or across multiple tones. The tones  409 - 1  through  409 -n correspond to the tones  317 - 1  through  317 -n of  FIG. 3 . Each tone  409 - 1  through  409 -n includes a symbol encoded in a circular signal space constellation. 
   Each of the tones  409 - 1  through  409 -n are supplied to constellation decoder  412  which outputs an index representing the constellation point received for each of the tones. This serial group of indices is supplied via connection  414  to optional fractional decoder  416 . The optional fractional decoder  416  converts the groups of indices, or words, into a new set of binary words, which are restored in their original order. Trellis decoder  411  communicates with constellation decoder  412  via connection  407 . 
   The optional fractional decoder  416  supplies the serial words via connection  418  to bit ordering element  421 . If the fractional decoder  416  is omitted, the constellation decoder  412  supplies the serial words on connection  414  directly to the bit ordering element  421 . The bit ordering element  421  orders the bits into their original order and supplies the serial stream on connection  422  to RS de-interleaver  424 , which de-interleaves the Reed-Solomon encoded bit stream. The de-interleaved bit stream is supplied on connection  426  to RS decoder  427 . RS decoder  427  removes the Reed-Solomon forward error correction and supplies the scrambled bit stream via connection  428  to descrambler  429 . Descrambler  429  descrambles the serial bit stream and supplies the original bit stream via connection  207  through the input/output element  202  ( FIG. 2 ) to DTE  101 . 
     FIG. 6A  is a graph  600  illustrating a conventional DMT transmit spectrum. The vertical axis of graph  600  represents power and the horizontal axis represents frequency. The tones  601 - 1  through  601 -n illustrate that each tone in a DMT communication scheme is independently variable and adjustable depending on line conditions. Each tone  601 - 1  through  601 -n represents the power at that tone frequency. Further, tone  601 - 7  represents the presence of a bridged tap at frequency f 1 . The bridged tap at frequency f 1  prevents the tone  601 - 7  from transmitting significant energy and also negatively affects the tones  601 - 6  and  601 - 8 . 
     FIG. 6B  is a graph  610  illustrating a DMT transmit spectrum after operation of the power and spectrum control feedback loop of  FIG. 3 . The graph  610  illustrates that each of the tones  611 - 1  through  611 -n is equalized, thus compensating the bridged tap represented by tone  601 - 7  in  FIG. 6A  and maintaining the specified PSD, which is a flat spectrum for the case illustrated in  FIG. 6B . 
   While various embodiments of the invention have been described, it will be apparent to those of ordinary skill in the art that many more embodiments and implementations are possible that are within the scope of this invention. Accordingly, the invention is not to be restricted except in light of the attached claims and their equivalents.