Abstract:
Disclosed is a control circuit for control of a semiconductor switching device, such as an IGBT. The control circuit comprising a first feedback path between a first electrode and a control electrode of said semiconductor switching device which has a capacitance. The circuit is operable such that the capacitance in the first feedback path is dependent on the voltage level at said first electrode. In another embodiment the control circuit is operable such that a feedback signal begins to flow in the first feedback path immediately as the semiconductor switching device begins switching off, thereby causing a control action on the semiconductor switching device.

Description:
[0001]    The present disclosure relates to control of semiconductor switching devices, and in particular insulated-gate bipolar transistors (IGBT). 
       BACKGROUND 
       [0002]    The state of the art in power converters provides an adjustable voltage and frequency to the output through a pulse width modulated (PWM, pulse width modulation) voltage source inverter drive. Power converters can be used in uninterruptible power supplies (UPS), electric motors, etc. The PWM command is used in the power converter for controlling power to inertial electrical devices, made practical by modern electronic power switches. The duty cycle of a switch (ratio of on-time to total cycle time) is varied to achieve a desired average output voltage, current etc., when averaged over time. 
         [0003]    A typical power converter is a switching power converter. It has two or more power semiconductor devices such as power semiconductor switches. The power semiconductor switches can, for example, be implemented by insulated-gate bipolar transistors (IGBT). As the switching speed of such semiconductor switches increase and currents increase, it becomes increasingly difficult to limit turn-off voltages to a permissible range. Should the permissible voltage across the controlled path of a semiconductor switch be exceeded, it will be destroyed. A particularly critical case is that of a short circuit, where the rate of current change di/dt induces in the stray inductances a voltage which is added to the voltage present in any case. The resultant overvoltage can exceed the permissible voltages, particularly across the controlled path of the semiconductor switch. 
         [0004]    One approach to addressing the overvoltage is to increase the size of the resistor at the gate of the semiconductor switch. However, in order for this to be effective, the size of the resistor becomes too great and switching losses become unacceptable. Another method is to feedback the collector-emitter voltage to the gate of the semiconductor switch in order to maintain it in the on-state for a time determined by stray inductances in the high power circuit. The rise in the gate voltage limits the rise in the collector-emitter voltage. However, this method is largely ineffective due to the significant delay between the gate voltage falling below the Miller plateau and the rise of the collector-emitter voltage. 
         [0005]    Another approach is to provide an active clamp, such as described in U.S. Pat. No. 7,119,586. Here, the active clamp is incorporated in the circuit between the semiconductor switch&#39;s collector and the input to the gate driver stage. This active clamp determines the voltage across the emitter-collector path of the semiconductor switch, and in this manner detects the beginning of the cut-off state, thereby freezing the instantaneous value of the switching signal. Since the semiconductor switch remains longer at the voltage level of the Miller Plateau, a small rate of current change di/dt of the collector current is achieved when turning-off particularly high voltages. This method of active clamping tends to result in a change in collector current slope without significantly increasing switching losses. 
         [0006]    It is an aim of the present invention to address one or more of these issues with the prior art. 
         [0007]    In a first aspect of the invention there is provided a control circuit for control of a semiconductor switching device comprising a first feedback path between a first electrode and a control electrode of said semiconductor switching device, said first feedback path comprising a capacitance, said control circuit being operable such that the capacitance in the first feedback path is dependent on the voltage level at said first electrode. 
         [0008]    In a second aspect of the invention there is provided a control circuit for control of a semiconductor switching device comprising a first feedback path between a first electrode and a control electrode of said semiconductor switching device and comprising a capacitance, said control circuit being operable such that a feedback signal begins to flow in the first feedback path immediately as the semiconductor switching device begins switching off, thereby causing a control action on said semiconductor switching device. 
         [0009]    Other optional aspects are as disclosed in the appended dependent claims. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0010]    Embodiments of the invention will now be described, by way of example only, by reference to the accompanying drawings, in which: 
           [0011]      FIG. 1  shows an active driver circuit according to the prior art, driving an IGBT. 
           [0012]      FIG. 2  shows an active driver circuit according to the present invention, driving an IGBT; 
           [0013]      FIG. 3  shows traces of a number of signal levels over time for an IGBT without any active voltage control; and 
           [0014]      FIGS. 4 and 5  show traces of a number of signal levels over time for an IGBT driven using an active driver circuit according to an embodiment of the invention. 
       
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
       [0015]      FIG. 1  shows a prior art active driver circuit driving an Insulated Gate Bipolar Transistor (IGBT)  100 . The driving circuit receives a switching signal such as a pulse width modulated signal PWM, and drives the IGBT  100  in dependence on the switching signal PWM. The driving circuit consists of at least one driver stage  110 , which consists, in this example, of a push-pull emitter follower comprising two bipolar transistors  120   a  and  120   b.  One or more further driving stages may precede driving stage  110 . The output of the driver stage  110  is connected to the gate of IGBT  100  via gate resistor R G . A first feedback path between collector and the input of driver stage  110  is provided. This first feedback path comprises transient-voltage-suppression (TVS) diode  130 , capacitor  170 , resistor  180  and switch  190 . Also provided is a second feedback path between collector and gate of the IGBT. This path comprises TVS diode  130 , further TVS diodes in series  140 , diode  150  and resistor  160 . 
         [0016]    This circuit operates as follows. The PWM signal is received via the driver stage  110  at the IGBT&#39;s  100  gate. As the gate voltage changes polarity, the IGBT  100  is cut off and the voltage between emitter and collector of the IGBT  100  rises. Once the threshold set by TVS diode  130  is reached the potential at the capacitor  170  changes and a current flows through the first feedback path via the resistor  180 . The voltage dropped across the resistor  180  switches on the switch  190 , causing a current to flow through the switch  190 . Due to this current, in turn, a positive voltage is dropped across the resistance  195  which holds the IGBT  100  at its Miller Plateau directly via the driver stage  110 . Consequently, the IGBT  100  remains in its active region. This significantly reduces the current slope, and hence the instantaneous rate of change in voltage dv/dt across the IGBT is limited. 
         [0017]    The second, optional, feedback path provides a further clamping action on the IGBT&#39;s gate by directly applying the collector-emitter current to the gate, via resistance  160  and TVS diodes  130 ,  140 . The TVS diodes  130 , 140  in series result in a higher threshold being required before this further clamping action takes effect, compared to the threshold required for the clamping action of the first feedback circuit to take effect. This only happens when in an overvoltage situation, under normal operation the collector-emitter voltage will not exceed this higher threshold. 
         [0018]      FIG. 2  shows a modified active driver circuit driving an IGBT  200 . The circuit operates in a similar manner to that of  FIG. 1 . The main difference is in the first feedback path, where there is a plurality (in this example, three) capacitors  270   a,    270   b,    270   c,  in parallel. The first capacitor  270   a  has no TVS diode at its input, and therefore the rising collector-emitter voltage is immediately applied to it, resulting in a current flow in the first feedback path. However, as the collector-emitter voltage continues to increase, the threshold set by TVS diode  230   a  is met, followed by the threshold set by the combination of TVS diodes  230   a  and  230   b.  This results in, firstly, current being conducted through capacitor  270   b  in parallel with  270   a  and then through all three capacitors  270   a,    270   b,    270   c.  The instantaneous current i control1  through this feedback path is given by: 
         [0000]    
       
      
       i 
       control1 
       =C·dv 
       ce 
       /dt  
      
     
         [0000]    where C is the capacitance in the feedback loop and dv ce /dt is the instantaneous rate of change of the IGBT&#39;s  200  collector-emitter voltage. Consequently, as each threshold is passed, the effective capacitance in the feedback loop increases, and with it the current in the feedback loop. 
         [0019]    The way that the feedback current i control1  is used to control the driver stage  210  also differs from the circuit of  FIG. 1 .  FIG. 2  shows a core control command CCC (which causes the PWM signal) input via isolation  205  and amplification  215  to the driver stage  210 . This core control command also provides an inhibition command  225  at the base of switch T 2 . In parallel with switch T 2  is resistor R 2  and diode D 3 . The first feedback path is connected to the gate of switch T 1 , which connects the positive rail to the driver stage  210  input via diode D 2 . 
         [0020]    Feedback current i control1  passes through resistor R 1  with a consequent current being injected into resistor R 2 . The voltage across resistor R 2  rises, and switch T 1  is turned on. Switch T 1  acts as a current amplifier in order to control transistor  220   a  such that the power IGBT  200  stays in its active region as described in relation to  FIG. 1 . 
         [0021]    This action is inhibited as a result of the inhibition command  225  after a delay of (in this example) 1 μs. After that time, switch T 2  is turned on and no more action from resistor R 2  can occur. The capacitors  270   a    270   b    270   c  are discharged through diode D 3  and resistor R 1  when the power IGBT  200  is turned on, and are therefore ready to control the next turn off of the power IGBT  200 . 
         [0022]    The second feedback path operates in the same was as that of  FIG. 1 . 
         [0023]    A main difference with the circuit of  FIG. 1  is that the feedback current i control1  begins to flow in the first feedback path, and therefore the control action occurs, immediately on the IGBT being switched off. In addition, the control signal in the first feedback loop increases with the collector emitter current. In this way, the multilevel thresholds lead to a better overvoltage control, without a significant amount of extra losses. 
         [0024]      FIG. 3  shows traces for the collector current Ic, the gate voltage Vge, the collector-emitter overvoltage Vce, the energy losses e loss  and the gate current Ig for a power IGBT without active voltage control, while  FIG. 4  shows the same traces obtained for a power IGBT using the circuit of  FIG. 2 . 
         [0025]    The IGBT is turning off 400 A in both of these Figures. In  FIG. 4 , the gate voltage Vge can be seen to have a number of peaks, each representing a threshold reached. This results in better control of the collector-emitter overvoltage Vce. In  FIG. 3  the overvoltage can be seen to peak above 700V while in  FIG. 4  it is constrained below 600V. 
         [0026]      FIG. 5  shows the same traces as  FIG. 4  using the circuit of  FIG. 2 , but this time turning off 800 A. The IGBT has a max current of 400 A (this is a one shot test, as the IGBT cannot withstand this current permanently). At this current, it can be seen that the peak overvoltage Vice is very close to (only very slightly higher) than that shown in  FIG. 4 , for which the IGBT switching off 400 A. 
         [0027]    While this detailed description has set forth some embodiments of the present invention, the appended claims cover other embodiments of the present invention which differ from the described embodiments according to various modifications and improvements and/or which can be envisaged without departing from the spirit or scope of the invention.