Abstract:
A receiver correlator structure for an ultra wide bandwidth communication system includes an antenna, a mixer, a bandpass filter, and a convertor. The receiver receives, via the antenna, an ultra wide bandwidth signal comprising a sequence of wavelets of particular shapes and positions, and transmits the received ultra wide bandwidth signal to the mixer. The mixer also receives and mixes with the received ultra wide bandwidth signal a local ultra wide bandwidth signal comprising a sequence of wavelets of particular shapes and positions correlated to the received ultra wide bandwidth signal. The bandpass filter removes the DC components from the mixed signal, and provides the resultant signal to the convertor. The receiver structure eliminates the local ultra wide bandwidth signal AC bias and DC bias terms and 1/f noise, yet detects long sequences of logical 1&#39;s and 0&#39;s, and allows operation with reduced bandwidth convertors.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
     The present application is a continuation application of U.S. patent application Ser. No. 09/685,200, filed Oct. 10, 2000, entitled LEAKAGE NULLING RECEIVER CORRELATOR STRUCTURE AND METHOD FOR ULTRA WIDE BANDWIDTH COMMUNICATION SYSTEM, now issued as U.S. Pat. No. 6,937,646. 
    
    
     The present document contains subject matter related to that disclosed in the following commonly owned, and co-pending U.S. patent application Ser. No. 09/209,460 filed Dec. 11, 1998, entitled ULTRA WIDE BANDWIDTH SPREAD-SPECTRUM COMMUNICATIONS SYSTEM, now issued as U.S. Pat. No. 6,700,939; Ser. No. 09/633,815 filed Aug. 7, 2000, entitled ELECTRICALLY SMALL PLANAR UWB ANTENNA; Ser. No. 09/563,292 filed May 3, 2000, entitled PLANAR ULTRA WIDE BAND ANTENNA WITH INTEGRATED ELECTRONICS, now issued as U.S. Pat. No. 6,351,246; Ser. No. 60/207,225 filed May 26, 2000, entitled ULTRA WIDEBAND COMMUNICATION SYSTEM AND METHOD; Ser. No. 09/685,198 filed Oct. 10, 2000, entitled ANALOG SIGNAL SEPARATOR FOR UWB VERSUS NARROWBAND SIGNALS, now issued as U.S. Pat. No. 7,006,553; Ser. No. 60/238,466 filed Oct. 10, 2000, entitled ULTRA WIDE BANDWIDTH NOISE CANCELLATION MECHANISM AND METHOD; Ser. No. 60/217,099 filed Jul. 10, 2000, entitled MULTIMEDIA WIRELESS PERSONAL AREA. NETWORK (WPAN) PHYSICAL LAYER SYSTEM AND METHOD; Ser. No. 09/685,203 filed Oct. 10, 2000, entitled SYSTEM AND METHOD FOR BASEBAND REMOVAL OF NARROWBAND INTERFERENCE IN ULTRA WIDEBAND SIGNALS, now issued as U.S. Pat. No. 6,834,073; Ser. No. 09/685,197 filed Oct. 10, 2000, entitled MODE CONTROLLER FOR SIGNAL ACQUISITION AND TRACKING IN AN ULTRA. WIDEBAND COMMUNICATION SYSTEM, now issued as U.S. Pat. No. 6,965,630; Ser. No. 09/684,400 filed Oct. 10, 2000, entitled ULTRA WIDEBAND COMMUNICATION SYSTEM, METHOD, AND DEVICE WITH LOW NOISE PULSE FORMATION, now issued as U.S. Pat. No. 6,735,238; Ser. No. 09/685,195 filed Oct. 10, 2000, entitled ULTRA WIDE BANDWIDTH SYSTEM AND METHOD FOR FAST SYNCHRONIZATION, now issued as U.S. Pat. No. 6,925,108; Ser. No. 09/684,401 filed Oct. 10, 2000, entitled ULTRA WIDE BANDWIDTH SYSTEM AND METHOD FOR FAST SYNCHRONIZATION USING SUB CODE SPINS, now issued as U.S. Pat. No. 6,967,993; Ser. No. 09/685,196 filed Oct. 10, 2000, entitled ULTRA WIDE BANDWIDTH SYSTEM AND METHOD FOR FAST SYNCHRONIZATION USING MULTIPLE DETECTION ARMS, now issued as U.S. Pat. No. 7,079,604; Ser. No. 09/685,202 filed Oct. 10, 2000, entitled METHOD AND SYSTEM FOR ENABLING DEVICE FUNCTIONS BASED ON DISTANCE INFORMATION, now issued as U.S. Pat. No. 7,058,414; Ser. No. 09/685,201 filed Oct. 10, 2000, entitled CARRIERLESS ULTRA WIDEBAND WIRELESS SIGNALS FOR CONVEYING APPLICATION DATA, now issued as U.S. Pat. No. 6,505,032; Ser. No. 09/685,205 filed Oct. 10, 2000, entitled SYSTEM AND METHOD FOR GENERATING ULTRA WIDEBAND PULSES, now issued as U.S. Pat. No. 7,010,056; Ser. No. 09/684,782 filed Oct. 10, 2000, entitled ULTRA WIDEBAND COMMUNICATION SYSTEM, METHOD, AND DEVICE WITH LOW NOISE RECEPTION, now issued as U.S. Pat. No. 6,859,506; and Ser. No. 09/685,199 filed Oct. 10, 2000, entitled A LOW POWER, HIGH RESOLUTION TIMING GENERATOR FOR ULTRA-WIDE BANDWIDTH COMMUNICATION SYSTEMS, now issued as U.S. Pat. No. 6,975,665, the entire contents of each of which being incorporated herein by reference. 
     The entire contents of Lathi, “Modern Digital and Analog Communications Systems,” Holt, Rinehart and Winston, 1998 is incorporated herein by reference. 
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention is directed to ultra wide bandwidth communication systems and, more particularly, to a receiver included in ultra wide-band communications systems. 
     2. Description of the Related Art 
     An ultra wide bandwidth spread-spectrum communications system is disclosed in U.S. Utility patent application Ser. No. 09/209,460 filed Dec. 11, 1998. In the ultra wide bandwidth spread-spectrum communication system disclosed in the foregoing application, data is transmitted between a transmitter and a receiver through multiple pathways. Transmission of data from a transmitter to a receiver through multiple pathways is referred to as multi-path communications. 
     The transmitted data is encoded by the transmitter using sequences of N wavelets of particular shapes and positions. Each of the sequences of N wavelets is referred to as an N-length code and represents one bit of digital data, either logical “1” or logical “0”. Whether a particular sequence of wavelets represents a logical “1” or a logical “0” is selected arbitrarily according to the source data at the transmitter. For example, a logical “1” would be represented by the sequence of wavelets shown in  FIG. 1A , while a logical “0” would be represented by a the same sequence of wavelets inverted as shown in  FIG. 1B . 
     As shown in  FIGS. 1A and 1B , the period of each wavelet of each waveform RF is 500 picoseconds (ps), with some arbitrary delay occurring between each wavelet referred to as a chip period. That is, a chip period refers to time from one wavelet to a corresponding location on the next wavelet, such as center-to-center. The N-length code is transmitted between t=t 0  and t=T. 
       FIG. 2  shows a receiver correlator structure  10  which receives data organized into N-length codes. The data, carried on signal RF, impinges on an antenna  12 , and is transmitted to coupled mixer  14 . Mixer  14  also receives a local oscillator signal, LO, generated by a wavelet generator (not shown in  FIG. 2 ). 
     Signal LO is a pulse stream divided into N-length codes, matched in time to the N-length codes of signal RF. Each N-length code of signal LO represents the same logical value as the prior four-length code of signal LO. That is, the signal LO provides the same value, either logical “0” or logical “1” code as sent by the transmitting source device, to mixer  14 . Mixer  14  mixes signal RR with signal LO to produce signal IF=RF*LO. 
     An example of a waveform for signal LO, having a four-length code corresponding to logical “0” code, is shown in  FIG. 1B  traveling between time t=t 0  and t=T. 
       FIG. 3  shows the simplest code in which every chip is the same. 
       FIGS. 4A and 4B  show example waveforms corresponding to signal IF at the output of mixer  14 . The resultant IF waveform shown in  FIG. 4A  is produced when signal RF, having a value of logical “1” code shown in  FIG. 1A , is mixed by mixer  14  with signal LO also having a value of logical “1” code. The resultant IF waveform shown in  FIG. 4B  is produced when signal REF having a value of logical “0” code shown in  FIG. 1B  is mixed by mixer  14  with signal LO having a value of logical “1” code. 
     Signal IF is then integrated by integrator  16  and, subsequently, converted to a digital signal D by analog-to-digital (A/D) converter  18 . Digital signal processor (DSP)  20  then removes noise from the resultant integrated digital signal by implementing algorithms known to one of ordinary skill in the art. 
     Mixer  14 , integrator  16 , analog-to-digital convertor  18  and digital signal processor  20  are conventional components. 
     Mixer  14  and integrator  16  comprise, ideally, a mathematically matched filter for signal RF. Moreover, integrator  16  is aligned correctly with each bit of data and as would be apparent to one of ordinary skill in the art, will either ramp up or ramp down, depending upon the value corresponding to a particular bit integrated by integrator  16 . 
     Since imperfect isolation occurs between signal LO and signal RF at mixer  14 , resulting in leakage current  22  between signal LO and signal RF, a bias in signal IF is included at the output of mixer  14 . 
       FIGS. 5A-5C  show graphs of relative values of signals IF, I, and D shown in  FIG. 2 . As shown in  FIGS. 5A-5C , if signal IF is as shown in  FIG. 4A , then the waveform of signal I would appear as shown in  FIG. 5B , and the waveform of signal D would appear as shown in  FIG. 5C . The waveform of signal I shown in  FIG. 5B  includes an upward slope from 0V to 1V, from 1V to 2V, etc. due to both DC and AC components present in signal I, which were introduced by signal LO into mixer  14  and integrated by integrator  16 . 
     At time t=t 0 , and at each subsequent sampling point through t=T executed by integrator  16 , the value of signal I increases by 1, but that increase is a sloped increase due to DC components included in I, until a value of “4” is reached at time t=T. Between t=t 0  and t=T, the value of D i  is 0. At time t=T, A/D  18  samples I and increases the value of D to D=4. That is, D i +1=4. Thereafter, the integrator  16  is reset to ground causing D to be reset to 0. That is, D i +2=0. 
     Referring again to  FIG. 2 , when signal LO is transmitted to mixer  14 , leakage current  22  results in a DC offset (or bias) being provided to signal IF at the output of mixer  14 , and the resultant DC offset is integrated by integrator  16 . Moreover, the value of signal LO is relatively large being typically in the range of volts) compared to the value of signal RF (being typically in the range of microvolts), making it difficult for receiver correlator structure  10  to distinguish between signal LO and signal RF after signals LO and RF are mixed by mixer  14 . 
     More particularly, signal LO interferes with signal RF by coupling to RF, by radiating into the air and thus impinging upon antenna  12 , and by radiating to and bouncing back from antenna  12 . 
     Moreover, signal LO when provided to mixer  14  causes the above-mentioned bias to change randomly over time, resulting in mixer  14  having an output quantity of noise proportional to 1/f, where f is a frequency. 
     A problem with the ultra wide bandwidth receivers of the related art is that the mixer may transmit a leakage current from the local oscillator signal (which is typically in the range of microvolts) to the input of the mixer which receives the pulses, resulting in a DC offset (or bias) from the leakage current being provided at the output of the mixer. This bias then propagates as noise through the rest of the receiver, and interferes with the decoding of the information carried on the pulses. 
     Another problem with ultra wide bandwidth receivers of the related art is that the analog-to-digital converter may not sample the waveform output by the bandpass receiver at a sample point which corresponds to the maximum height of the waveform. 
     SUMMARY OF THE INVENTION 
     It is an object of the present invention to eliminate the DC component of signal IF yet optimally detect a long stream of logical “1”&#39;s or “0”&#39;s while using real, non-ideal components such as diode ring mixers, FET bridge mixers, Gilbert Cell mixers, dual-gate MOS FETs, etc. 
     It is another object to the present invention to reset the bias introduced to a received signal by a mixer in a receiver correlator to 0 in a single chip period. 
     It is also an object of the invention to null the AC-bias signal caused by leakage between signal LO and signal IF at the sampling instant, and to do so without the need for digital estimation and subtraction algorithms and processing. 
     It is a further object of the invention to broaden the peak of the waveform at the output of the bandpass filter so that a converter slower than an ultra wide bandwidth converter can be included in the receiver correlator of the present invention. 
     It is another object of the invention to eliminate 1/f noise from all mixer leakage terms and mixing terms and non-linearities. 
     The above objects can be attained by a system that receives a sequence of wavelets that are not back to back but have sufficient spacing T S  as input signal RF and as a local oscillator signal LO. 
     More particularly, the present invention is a receiver correlator structure and corresponding method. The receiver correlator structure of the present invention comprises an antenna, a mixer, a bandpass filter, and a converter. The antenna receives an ultra wide bandwidth signal comprising a sequence of wavelets of particular shapes and positions. The mixer is coupled to the antenna and receives from the antenna the ultra wide bandwidth signal and mixing the ultra wide bandwidth signal with a local ultra wide bandwidth signal comprising a sequence of wavelets of particular shapes and positions correlated to the sequence of wavelets of particular shapes and positions of the received ultra wide bandwidth signal. The bandpass filter is coupled to the mixer output and receives the mixed ultra wide bandwidth signal, removing the DC components therefrom. The bandpass filter outputs a resultant signal in which an initial peak of the resultant signal is proportional to energy included in the mixed ultra wide bandwidth signal and post signal decay of the resultant signal to zero occurs in T S  time. The convertor is coupled to the bandpass filter and converts the resultant signal at the initial peak to a digital output signal. The convertor is coupled optionally to a digital signal processor. The converter is one of an analog-to-digital converter, a comparator, or a sample and hold circuit coupled in series to an integrator then to an analog-to-digital converter. 
     These together with other objects and advantages which will be subsequently apparent, reside in the details of construction and operation as more fully hereinafter described and claimed, reference being had to the accompanying drawings forming a part hereof, wherein like numerals refer to like parts throughout. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1A  shows a sequence of wavelets (or chips) representing a logical “1” code. 
         FIG. 1B  shows a sequence of wavelets (chips) representing a logical “0” code. 
         FIG. 2  shows a receiver correlator structure 
         FIG. 3  shows an example of a waveform output by a local oscillator LO and input to the receiver correlator structure shown in  FIG. 2 . 
         FIGS. 4A and 4B  show example waveforms of signal IF shown in  FIG. 2 . 
         FIGS. 5A ,  5 B, and  5 C show graphs of relative values of signals IF, I, and D shown in  FIG. 2 . 
         FIG. 6  shows a receiver correlator of the present invention. 
         FIG. 7  shows an example of a waveform presented by local oscillator LO to the receiver correlator of the present invention. 
         FIG. 8  shows a series of signals RF input to the receiver correlator of the present invention shown in  FIG. 6 . 
         FIG. 9A  shows a detailed diagram of the receiver correlator of the present invention. 
         FIG. 9B  shows another embodiment of the receiver correlator of the present invention. 
         FIGS. 9C ,  9 D, and  9 E show alternate embodiments of converter  44 . 
         FIGS. 10A and 11B  show examples of waveform IF at the output of mixer  40  of the receiver correlator of the present invention. 
         FIG. 11A  shows band pass filter  42  being applied to signal IF. 
         FIG. 11B  shows the resultant waveform at the output of band pass filter  42 . 
         FIG. 12  shows an example of a series of wavelets transmitting data to the receiver correlator structure of the present invention. 
         FIG. 13A  shows one embodiment of band pass filter  42 . 
         FIG. 13B  shows another embodiment of band pass filter  42 . 
         FIG. 14  shows a table of outputs by receiver correlator  10  of the related art and receiver correlator  30  of the present invention. 
         FIG. 15  shows an ultra wide bandwidth communication system in which the receiver correlator of the present invention would be included. 
         FIG. 16  shows a diagram of a UWB Waveform Correlator of the ultra wide bandwidth communication system shown in  FIG. 15 . 
         FIG. 17  shows a diagram of a Radio Controller and Interface of the ultra wide bandwidth communication system shown in  FIG. 15 . 
     
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       FIG. 6  shows a receiver correlator  30  of the present invention which receives from transmitter  32  an ultra wide bandwidth signal RF. Ultra wide bandwidth signal RF comprises a sequence of wavelets of particular shapes and positions. Transmitter  32  is disclosed in Ser. No. 09/685,205 filed Oct. 10, 2000, entitled SYSTEM AND METHOD FOR GENERATING ULTRA WIDEBAND PULSES, incorporated herein by reference, and in Ser. No. 09/684,400 filed Oct. 10, 2000, entitled ULTRA WIDEBAND COMMUNICATION SYSTEM, METHOD, AND DEVICE WITH LOW NOISE PULSE FORMATION, incorporated herein by reference. 
     In addition, the receiver correlator structure  30  of the present invention receives an ultra wide bandwidth signal LO. Ultra wide bandwidth signal LO comprises a sequence of wavelets of particular shapes and positions corresponding to ultra wide bandwidth signal RF. Ultra wide bandwidth signal LO is produced by timing generator  36  as disclosed in co-pending application Ser. No. 09/685,199 filed Oct. 10, 2000, entitled A LOW POWER, HIGH RESOLUTION TIMING GENERATOR FOR ULTRA-WIDE BANDWIDTH COMMUNICATION SYSTEMS, incorporated herein by reference. More particularly, timing generator  36 , based upon inputs of frequency, phase, and time dither, prompts wavelet generator  34  to generate signal LO, which is time-aligned with signal RF. 
     Wavelet generator  34  is disclosed in co-pending application Ser. No. 09/685,205 filed Oct. 10, 2000, entitled SYSTEM AND METHOD FOR GENERATING ULTRA WIDEBAND PULSES, incorporated herein by reference. 
     The shape of the wavelet of signal LO generated by wavelet generator  34  can vary, with shapes such as biphase, multi-amplitude, multi-phase, and chirp. 
       FIG. 7  shows an example of a waveform presented by signal LO in  FIG. 6 . The waveform LO shown in  FIG. 7  was generated by wavelet generator  34  and has a constant pulse shape. 
       FIG. 8  shows a series of signals RF 1 , RF 2 , RF 3 , . . . which are wavelets, having a period of T S . Each wavelet, or series of wavelets, of ultra wide bandwidth signal RF corresponds to a logical “1” or a logical “0”, depending upon the shape of the wavelet of signal RF. That is, RF 1  of  FIG. 8  could correspond to logical “1” (and signal LO of  FIG. 7 ), while RF 2  and RF 3  of  FIG. 8  could each correspond to logical “0”. As shown in  FIG. 8 , T p  is less than T S  (the center-to-enter clock period), and T pp  (peak to peak) is less than T p . T p  is the width of the wavelet, and is typically twice T pp . An example of T pp  would be 330 pico-seconds (peak-to-peak); an example of T S  would be 20 nano-seconds. 
       FIG. 9A  shows a more detailed diagram of receiver correlator structure  30  of the present invention. As shown in  FIG. 9A , signal RF impinges upon antenna  38 , and is coupled to mixer  40 . Antenna  38  could be a conventional antenna or could be of the structure disclosed in co-pending U.S. patent application Ser. No. 09/563,292 filed May 3, 2000, entitled PLANAR ULTRA WIDE BAND ANTENNA WITH INTEGRATED ELECTRONICS, incorporated herein by reference. 
     Mixer  40  receives signal RF and local oscillator signal LO, which is time-aligned with signal RF as discussed in co-pending U.S. patent application Ser. No. 09/685,197 filed Oct. 10, 2000, entitled MODE CONTROLLER FOR SIGNAL ACQUISITION AND TRACKING IN AN ULTRA WIDEBAND COMMUNICATION SYSTEM, incorporated herein by reference, and in Ser. No. 09/685,195 filed Oct. 10, 2000, entitled ULTRA WIDE BANDWIDTH SYSTEM AND METHOD FOR FAST SYNCHRONIZATION, incorporated herein by reference. 
     In one embodiment, the pulse shape of signal LO can be constant. That is, each waveform generated by wavelet generator  34  (shown in  FIG. 6 ) is the same as the prior waveform of signal LO, and corresponds to either a logical “1” or a logical “0”, but does not vary from pulse to pulse. Consequently, leakage L (shown in  FIGS. 9A and 9B ) of signal LO between mixer  40  and antenna  38  is always the same from pulse to pulse of signal LO. This leakage L is non-linear, and is removed as discussed herein below. 
     Mixer  40  is, for example, a Mini-circuits ADE-42 MH. 
     After signals RF and LO are mixed by mixer  40 , the resultant, mixed signal IF is transmitted from mixer  40  to band pass filter  42 . Band pass filter  42  removes DC components from signal IF. That is, band pass filter  42  removes any bias introduced by mixer  40  by resetting the bias to 0 in a single chip period. Thus, in band pass filter  42 , the settling time is less then the bit. More particularly, band pass filter  42  removes 1/f noise. Band pass filter  42  is discussed in further detail with reference to  FIGS. 13A and 13B . 
       FIG. 9B  shows another embodiment of the present invention which includes adder  43 . Adder  43  has a bias voltage V B . V B  is set such that when signal RF=0, the output of convertor  44  (if convertor  44  is an analog-to-digital convertor) is OV at sampling point S. If converter  44  is a comparator, then. V B  is set such that the comparator has a 50% duty cycle. If V B  is not set in this way, then DSP  46  would have to estimate V B  across a series of bits decoded from the wavelets of received ultra wide bandwidth signal RF. However, setting V B  as described would save time and battery life over having DSP  46  estimate V B . 
       FIGS. 9C ,  9 D, and  9 E show alternate embodiments of the convertor  44 .  FIG. 9C  shows analog-to-digital converter  92 ;  FIG. 9D  shows comparator  94 ; and  FIG. 9E  shows sample-and-hold circuit  96  coupled in series to integrator  98 , which is coupled in series to analog-to-digital convertor  99 . 
     In one embodiment, integrator  98  would function as a D-latch circuit, in which the integrator would sample a signal input thereto on the rising edge of a clock pulse input to the integrator  98 , and transfers to the output of integrator  98  the signals that were integrated since the prior clock pulse. Such an integrator  98  would comprise a ping-pong circuit, which would comprise two integrators functioning during alternate clock periods. An example of such an integrator  98  is disclosed in U.S. patent application Ser. No. 09/209,460 filed Dec. 11, 1998, entitled ULTRA WIDE BANDWIDTH SPREAD-SPECTRUM COMMUNICATIONS SYSTEM, incorporated herein by reference, and in U.S. patent application Ser. No. 09/633,815 filed Aug. 7, 2000, entitled ELECTRICALLY SMALL PLANAR UWB ANTENNA, incorporated herein by reference. 
     An example of waveform IF, at the output of mixer  40 , is shown in  FIG. 10A . For the shape of waveform IF shown in  FIG. 10A , the shape of the waveform input in signal RF corresponds to a logical “1”, and the shape of the waveform input by signal LO also corresponds to a logical “1”. Of course, if RF and LO were logically opposite to each other, then the shape of waveform IF at the output of mixer  40  would correspond to waveform IF shown in  FIG. 10B . 
     Moreover, in the signal IF shown in  FIGS. 10A and 10B , the peak-to-peak difference between pulses is typically less than 1 nanosecond, though in some applications could be longer. 
     Band pass filter  42  accommodates waveform IF as shown in  FIG. 10A  and in  FIG. 10B . That is, band pass filter  42  recognizes dynamically whether signal IF corresponds to the waveform shown in  FIG. 10A  or corresponds to the waveform shown in  FIG. 10B , and removes the DC components, and therefore the bias introduced by mixer  40 , from either. 
       FIG. 11A  shows band pass filter  42  being applied to signal IF, and  FIG. 11B  shows resultant signal B, at the output of the band pass filter  42 , after band pass filter  42  is applied to signal IF. That is, signal B corresponds to signal IF, but with DC components present in signal IF removed from signal B by band pass filter  42 , and the AC signal leaked by mixer  40  from signal LO, removed at the sampling instant S. Also as shown in  FIG. 11A , in the receiver correlator of the present invention, the sampling instant S by the convertor  44  occurs at the relative maximum (or peak) or relative minimum of the output waveform from the bandpass filter  42 . 
     As shown in  FIG. 11A , the settling time of the band pass filter  42  is less than the bit period of the data encoded in signal IF. 
     Signal B is then transmitted to convertor  44 , which samples signal B at each respective peak thereof. That is, convertor  44  samples signal B at sampling points S shown in  FIG. 11A . Convertor  44  is an analog-to-digital convertor. Alternatively, convertor  44  is a comparator. Alternatively still, convertor  44  is a sample-and-hold circuit coupled in series to an integrator and to an analog-to-digital convertor. 
     Convertor  44  is biased by V b  such that the output of the convertor  44  is at the 0-volt threshold (or zero mean), when there is no RF signal or the RF signal is white Gaussian noise. That is, each bit is resolved to 0 by the combination of band pass filter  42  and convertor  44 . The Convertor  44  is, for example a Texas Instruments TLV 5580. 
     For pulses (or wavelets) transmitted as signal RF which correspond to logical “1” or logical “0”, then output signal OF is output by receiver correlator structure  30  directly from Convertor  44 . 
     Each pulse transmitted as signal RF, and ultimately, as signal IF and signal B, corresponds to a logical “1” or a logical “0” in the embodiment described here and above. 
     Alternatively, a series of single spike pulses, RF, represent a bit. That is, a series of two consecutive single spike pulse correspond to either logical “1” or logical “0”. If a series of single spike pulses corresponds to either logical “1” or logical “0” then a digital signal processor DSP  46  is placed at the output of Convertor  44 . The DSP  46  receives the output signal from Convertor  44  and digitally integrates the output signal. That is, DSP  44  implements an algorithm readily apparent to those of skill in the art to sum or add the series of pulses such that each series of pulses is determined to be to correspond to either a logical “1” or a logical “0”. DSP  46  is, for example an Intel Pentium™. processor. 
       FIG. 12  shows a series of pulses, RF, which when received by receiver correlator structure  30  of the present invention, resolve through the activities of antenna  38 , mixer  40 , band pass filter  42 , Convertor  44  and DSP  46  as described herein above, to a logical “1”. Although the series of pulses RP correspond to logical “1” each single spike pulse is spaced from another single spike pulse by a period of T S . 
       FIG. 13A  shows an embodiment of band pass filter  42 . Referring now to  FIG. 13A , signal IF enters band pass filter  42  along an input port which is tied to ground through a 9-picoFarad capacitor  50  and a 51-ohm resister  52  and proceeds to 22-nanoHenry inductor  54 . The output of inductor  54  is tied to ground through 51-OHM resistor  56  and 100-nanoHenry inductor  58  and to 47-picoFarad capacitor  60 , which is input to amplifier ERA-5SM  62 . The output of amplifier  62  is tied to ground through 4-picoFarad capacitor  64  and to 22-nanoHenry inductor  66 . The output of inductor  66  is tied, concurrently, to ground through 4-picoFarad capacitor  68 , and to a 12-volt power source through 110-ohm resistor  70 , which is in parallel with 0.01-microFarad capacitor  72 , also tied to ground. The output of inductor  66  travels through 0.01-microFarad capacitor  74  to become output B. 
     An alternate embodiment of band pass filter  42  is shown in  FIG. 13B . Referring now to  FIG. 13B , input signal IF travels through capacitor  80 , the output of which is tied in parallel to capacitor  82 , capacitor  84  and ground through capacitor  86 . Signal IF continues through capacitor  84  into amplifier  86  and through capacitor  82  to the output of amplifier  86 . The output of amplifier  86  is tied through resistor  88  back to the input of amplifier  86  and is output from band pass filter  42  at signal B. 
       FIG. 14  shows a table  90  of outputs by receiver correlator  30  and receiver correlator  10 , based upon input signal RF and local oscillator LO. As shown in  FIG. 14 , when input signal RF is corresponds to logical “1” and local oscillator corresponds to logical “1”, receiver correlator  30  of the present invention outputs a value of +1, whereas receiver correlator  10  would output a value of 1+a, which is perhaps a value other than +1. Likewise, if RF corresponds to logical “0” and LO corresponds to logical “0”, receiver correlator  30  would output +1 whereas receiver correlator  10  would output a value of 1-b, which is also perhaps a value other than +1. In addition, if RF corresponds to logical “1”, and LO corresponds to logical “0” receiver correlator  30  outputs −1 and receiver correlator  10  outputs −1-b, which is perhaps a value other than −1. Lastly if RF corresponds to “0” and LO corresponds to “1” receiver correlator  30  outputs −1 and receiver correlator  10  outputs −1-a, which is also perhaps a value other than −1. 
     Receiver correlator  10  outputs a and b because of the affect of the bias between mixer  14  and antenna  12 . Moreover, a and b have 1/f noise that obscures the signal of interest. 
     The present invention has been described with, respect to the above-mentioned components. However, implementation of the above-mentioned functions is not limited in the present invention to the components described. For example, the present invention may be implemented in silicon on a signal chip or on multiple chips. 
     An ultra wide bandwidth system (that is, an ultra-wide band transceiver) in which the receiver correlator structure of the present invention could be implemented is shown in  FIG. 15 . The transceiver  100  shown in  FIG. 15  includes a receiver  112 , a transmitter  114 , and a Radio Controller and Interface  110 . The Receiver  112  includes antenna  102 , Front End  104 , UWB Waveform Correlator  106 , and Timing Generator  108 . 
     The Transmitter  114  includes antenna  116 , UWB Waveform Generator  118 , Timing Generator  108 , and Encoder  122 . 
     The Transceiver  100  is described in further detail in U.S. application Ser. No. 09/685,199 filed Oct. 10, 2000, entitled A LOW POWER, HIGH RESOLUTION TIMING GENERATOR FOR ULTRA-WIDE BANDWIDTH COMMUNICATION SYSTEMS, incorporated herein by reference. 
     The receiver correlator structure  30  of the present invention is incorporated into transceiver  100  as shown in  FIGS. 16 and 17 . That is, the mixer  40  and the bandpass filter  42  are included (along with wavelet generator  34 ) into UWB waveform correlator  106 , and the convertor  44  and the DSP  46  are included in the Radio Controller and Interface  110 . 
     That is an ultra wide bandwidth communication system of the present invention comprises a transmitter transmitting an ultra wide bandwidth signal comprising a sequence of wavelets of particular shapes and positions, a wavelet generator generating a local ultra wide bandwidth signal comprising a sequence of wavelets of particular shapes and positions corresponding to the transmitted ultra wide bandwidth signal, and a receiver correlator structure. The receiver correlator structure comprises an antenna receiving the transmitted ultra wide bandwidth signal, a mixer, coupled to the antenna, receiving from the antenna the received ultra wide bandwidth signal and mixing the ultra wide bandwidth signal with the local ultra wide bandwidth, signal, a bandpass filter, coupled to the mixer output, receiving the mixed ultra wide bandwidth signal, removing the DC components therefrom, and outputting a resultant signal, wherein an initial peak of the resultant signal is proportional to energy included in the mixed ultra wide bandwidth signal and post signal decay of the resultant signal to zero occurs in T S  time, and a convertor, coupled to the bandpass filter, converting the resultant signal at the initial peak to a digital output signal. 
     The many features and advantages of the invention are apparent from the detailed specification and, thus, it is intended by the appended claims to cover all such features and advantages of the invention which fall within the true spirit and scope of the invention. Further, since numerous modifications and changes will readily occur to those skilled in the art, it is not desired to limit the invention to the exact construction and operation illustrated and described, and accordingly all suitable modifications and equivalents may be resorted to, falling within the scope of the invention.