Abstract:
A system and method for simultaneously receiving first and second ensembles. The first ensemble includes a first signal from a first satellite, a first signal from a second satellite and a first signal from a terrestrial repeater. Likewise, the second ensemble includes a second signal from the first satellite, a second signal from the second satellite and a second signal from the terrestrial repeater. The inventive receiver further includes a mechanism for selectively outputting signals transmitted within the first and second ensembles. In the illustrative embodiment, the first signal from the second satellite is identical to the first signal from the first satellite. Similarly, the first signal from the terrestrial repeater is identical to the first signal from the first satellite.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to communications systems. More specifically, the present invention relates to satellite digital audio service (SDARS) receiver architectures. 
     While the present invention is described herein with reference to illustrative embodiments for particular applications, it should be understood that the invention is not limited thereto. Those having ordinary skill in the art and access to the teachings provided herein will recognize additional modifications, applications, and embodiments within the scope thereof and additional fields in which the present invention would be of significant utility. 
     2. Description of the Related Art 
     Satellite radio operators will soon provide digital quality radio broadcast services covering the entire continental United States. These services intend to offer approximately 100 channels, of which nearly 50 channels will provide music with the remaining stations offering news, sports, talk and data channels. According to C. E. Unterberg, Towbin, satellite radio has the capability to revolutionize the radio industry, in the same manner that cable and satellite television revolutionized the television industry. 
     Satellite radio has the ability to improve terrestrial radio&#39;s potential by offering a better audio quality, greater coverage and fewer commercials. Accordingly, in October of 1997, the Federal Communications Commission (FCC) granted two national satellite radio broadcast licenses. The FCC allocated 25 megahertz (MHz) of the electromagnetic spectrum for satellite digital broadcasting, 12.5 MHz of which are owned by CD Radio and 12.5 MHz of which are owned by the assignee of the present application “XM Satellite Radio Inc.”. The FCC further mandated the development of interoperable receivers for satellite radio reception, i.e. receivers capable of processing signals from either CD Radio or XM Radio broadcasts. The system plan for each licensee presently includes transmission of substantially the same program content from two or more geosynchronous or geostationary satellites to both mobile and fixed receivers on the ground. In urban canyons and other high population density areas with limited line-of-sight (LOS) satellite coverage, terrestrial repeaters will broadcast the same program content in order to improve coverage reliability. Some mobile receivers will be capable of simultaneously receiving signals from two satellites and one terrestrial repeater for combined spatial, frequency and time diversity, which provides significant mitigation against multipath and blockage of the satellite signals. In accordance with XM Radio&#39;s unique scheme, the 12.5 MHz band will be split into 6 slots. Four slots will be used for satellite transmission. The remaining two slots will be used for terrestrial re-enforcement. 
     In accordance with the XM frequency plan, each of two geostationary Hughes 702 satellites will transmit identical or at least similar program content. The signals transmitted with QPSK modulation from each satellite (hereinafter satellite 1  and satellite 2 ) will be time interleaved to lower the short-term time correlation and to maximize the robustness of the signal. For reliable reception, the LOS signals transmitted from satellite 1  are received, reformatted to Multi-Carrier Modulation (MCM) and rebroadcast by non-line-of-sight (NLOS) terrestrial repeaters. The assigned 12.5 MHz bandwidth (hereinafter the “XM” band) is partitioned into two equal ensembles or program groups A and B. The use of two ensembles allows 4096 Mbits/s of total user data to be distributed across the available bandwidth. Each ensemble will be transmitted by each satellite on a separate radio frequency (RF) carrier. Each RF carrier supports up to 50 channels of music or data in Time Division Multiplex (TDM) format. With terrestrial repeaters transmitting an A and a B signal, six total slots are provided, each slot being centered at a different RF carrier frequency. The use of two ensembles also allows for the implementation of a novel frequency plan which affords improved isolation between the satellite signals and the terrestrial signal when the receiver is located near the terrestrial repeater. 
     In any event, with different content being provided on each ensemble and inasmuch as data will be transmitted along with music content on one or both ensembles, it is conceivable that a listener will may want to access content on both ensembles simultaneously. 
     Unfortunately, there is currently no efficient satellite radio receiver architecture capable of receiving two ensembles simultaneously. Accordingly, system designers must consider either replicating the data on both ensembles or replicating the tuner within the receiver. Both approaches are unacceptably costly. As a result, there is a need in the art for satellite radio receiver architecture capable of receiving two ensembles simultaneously which will not require a replication of the tuner nor a replication of the data broadcast channel on both ensembles. 
     In accordance with the CD Radio frequency plan, the 12.5 MHz band will be split into 3 slots. Two slots will be used for satellite transmission. The remaining slot will be used for terrestrial re-enforcement. In order to comply with the FCC mandate to develop interoperable receivers, conventional satellite receiver architectures will require replicate analog signal paths and filters to process the various signal bandwidths associated with each assignees frequency plan. As a result, interoperable receivers will have substantially larger size and higher cost than non-interoperable receivers. 
     Thus, in addition, there is a need in the art for a satellite radio receiver architecture which reduces the cost and size overhead associated with the interoperation of XM Radio and CD Radio. 
     SUMMARY OF THE INVENTION 
     The need in the art is addressed by the system and method of the present invention. The inventive system includes a receiver adapted to simultaneously receive first and second ensembles. The first ensemble includes a first signal from a first satellite, a first signal from a second satellite and a first signal from a terrestrial repeater. Likewise, the said second ensemble includes a second signal from the first satellite, a second signal from the second satellite and a second signal from the terrestrial repeater. The inventive receiver further includes a mechanism for selectively outputting signals transmitted within the first and second ensembles. 
     In the illustrative embodiment, the first signal from the second satellite has program content similar (if not identical) to the first signal from the first satellite. Similarly, the first signal from the terrestrial repeater has program content similar (if not identical) to the first signal from the first satellite but it is transmitted using Multi-Carrier Modulation. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is an illustrative implementation of a satellite digital audio service (SDARS) system architecture constructed in accordance with the teachings of the present invention. 
     FIG. 2 is a diagram which illustrates the system of FIG. 1 in greater detail. 
     FIG. 3 a  is a diagram which depicts a frequency plan for a two-satellite SDARS broadcast system utilizing the XM band in accordance with the present teachings. 
     FIG. 3 b  is a diagram which depicts the frequency plan of FIG. 3 a  centered at baseband. 
     FIG. 4 a  is a diagram which depicts the CD Radio frequency plan. 
     FIG. 4 b  is a diagram which depicts the CD Radio frequency plan of FIG. 4 a  centered at baseband. 
     FIG. 5 is a block diagram of an illustrative implementation of an SDARS receiver constructed in accordance with the teachings of the present invention. 
     FIG. 6 is a detailed view of a receiver capable of receiving a single ensemble only. 
     FIG. 7 is a block diagram of a first embodiment of an SDARS receiver of the present invention. 
     FIG. 8 is an alternative embodiment of the SDARS receiver of FIG.  7 . 
     FIG. 9 is a block diagram of second alternative embodiment of the SDARS receiver of the present invention. 
     FIG. 10 is a block diagram of an alternative preferred embodiment of an SDARS receiver incorporating the teachings of the present invention. 
     FIG. 11 is a diagram which illustrates the benefits of direct digital conversion. 
     FIG. 12 is a diagram showing an XM full waveform receiver adapted to receive audio and data simultaneously. 
    
    
     DESCRIPTION OF THE INVENTION 
     Illustrative embodiments and exemplary applications will now be described with reference to the accompanying drawings to disclose the advantageous teachings of the present invention. 
     An illustrative implementation of a satellite digital audio service (SDARS) system architecture is depicted in FIG.  1 . The system  10  includes first and second geostationary satellites  12  and  14  which transmit line-of-sight (LOS) signals to SDARS receivers located on the surface of the earth. The satellites provide for interleaving and spatial diversity. (Those skilled in the art will appreciate that in the alternative, the signals from the two satellites could be delayed to provide time diversity.) The system  10  further includes plural terrestrial repeaters  16  which receive and retransmit the satellite signals to facilitate reliable reception in geographic areas where LOS reception from the satellites is obscured by tall buildings, hills, tunnels and other obstructions. The signals transmitted by the satellites  12  and  14  and the repeaters  16  are received by SDARS receiver  20 . As depicted in FIG. 1, the receivers  20  may be located in automobiles, handheld or stationary units for home or office use. The SDARS receivers  20  are designed to receive one or both of the satellite signals and the signals from the terrestrial repeaters and combine or select one of the signals as the receiver output as discussed more fully below. 
     FIG. 2 is a diagram which illustrates the system  10  of FIG. 1 in greater detail with a single satellite and a single terrestrial repeater FIG. 2 shows a broadcast segment  22  and a terrestrial repeater segment  24 . In the preferred embodiment, an incoming bit stream is encoded into a time division multiplexed (TDM) signal using a coding scheme such as MPEG by an encoder  26  of conventional design. The TDM bit stream is upconverted to RF by a conventional quadrature phase-shift keyed (QPSK) modulator  28 . The upconverted TDM bit stream is then uplinked to the satellites  12  and  14  by an antenna  30 . Those skilled in the art will appreciate that the present invention is not limited to the broadcast segment shown. Other systems may be used to provide signals to the satellites without departing from the scope of the present teachings. 
     The satellites  12  and  14  act as bent pipes and retransmit the uplinked signal to terrestrial repeaters  18  and portable receivers  20 . As illustrated in FIG. 2, the terrestrial repeater includes a receiver demodulator  34 , a de-interleaver and reformatter  35 , a terrestrial waveform modulator  36  and a frequency translator and amplifier  38 . The receiver and demodulator  34  downconverts the downlinked signal to a TDM bitstream. The de-interleaver and reformatter  35  reorders the TDM bitstream for the terrestrial waveform. The digital baseband signal is then applied to a terrestrial waveform modulator  36  (e.g. MCM or multiple carrier modulator) and then frequency translated to a carrier frequency prior to transmission. 
     As will be appreciated by those skilled in the art, the strength of the signal received close to the terrestrial repeaters will be higher than that received at a more distant location. A concern is that the terrestrial signal might interfere with the reception of the satellite signals by the receivers  30 . For this reason, in the best mode, a novel frequency plan such as that described below is utilized. 
     FIG. 3 a  is a diagram which depicts a frequency plan for a two-satellite SDARS broadcast system utilizing the XM band  40  in accordance with the present teachings. Each satellite transmits ensemble A and ensemble B. In accordance with the novel frequency plan of the present invention, two frequency slots  42  and  48  centered at frequencies  43  and  49  are assigned to the first satellite  12  and two frequency slots  44  and  46  centered at frequencies  45  and  47  are assigned to the second satellite  14 . In addition, two frequency slots  50  and  52  centered at frequencies  51  and  53  are assigned to the terrestrial repeaters  18 . Three frequency slots  42 ,  44  and  50  each carry identical program content assigned to ensemble A and the three frequency slots  48 ,  46  and  52  each carry identical program content assigned to ensemble B. As mentioned above, the repeaters  18  retransmit the signals received from satellite  12  as illustrated in FIG.  2 . 
     Returning to FIG. 3 a,  note that the frequency slots  42  and  48  associated with the satellite  12  are separated from the frequency slots  50  and  52  associated with the terrestrial repeaters  18  by the frequency slots  44  and  46  associated with satellite  14 . In this manner, any satellite interference created by a terrestrial repeater transmission will primarily impact only the signal from satellite  14  and not the signal from satellite  12 . As will be appreciated by those skilled in this art, this facilitates reliable reception by a receiver even while located in close proximity to a terrestrial repeater. 
     FIG. 4 a  is a diagram which depicts the CD Radio frequency plan and FIG. 4 b  is a diagram which depicts the CD Radio frequency plan of FIG. 4 a  centered at baseband. As depicted in FIGS. 4 a  and  4   b,  the three signals contain identical program content. The terrestrial signal is at the center of the band with the signals from the satellites on either side. 
     FIG. 5 is a block diagram of an illustrative implementation of an SDARS receiver  20  constructed in accordance with the teachings of the present invention. The receiver  20  includes an antenna module  100 , an RF tuner module  200 , a channel decoder  300 , a source decoder  400 , a digital control and status interface bus  600 , system controller  500 , data interface  700 , audio output circuit  800 , power supply  900 , and a user interface  1000 . 
     In order to appreciate the present teachings, reference is made to FIG.  6 . FIG. 6 is a detailed view of antenna module  100 ′ and tuner module  200 ′ capable of receiving a single ensemble only. In the preferred embodiment, the system disclosed in FIG. 6 is implemented in accordance with the teachings of U.S. patent application Ser. No. 09/435,317, entitled Tuner Architecture for Satellite and Terrestrial Reception of Signals, filed Nov. 14, 1999 by P. Marko and A. Nguyen, the teachings of which are incorporated herein by reference. The signal received by the antenna  110 ′ of the antenna module  100 ′ is amplified by a first low noise amplifier  122 ′ prior to being input to a first image filter  124 ′. The output of the first image filter  124 ′ is input to a second low noise amplifier  126 ′. The output of the second low noise amplifier  126 ′ is fed back to the first low noise amplifier  122 ′ via an automatic gain control (AGC) circuit  128 ′ for gain stabilization as will be appreciated by those skilled in the art. The output of the second low noise amplifier  126 ′ constitutes the output of the antenna module  100 ′ and is input to the tuner module  200 ′ via an RF cable  130 ′. 
     In the tuner module  200 ′, a second image filter  201 ′ receives the RF signal from the cable  130 ′ and provides an input to a third low noise amplifier  202 ′. The output of the third low noise amplifier  202 ′ is input to a first mixer  208 ′. The first mixer is driven by a dual resonator voltage controlled oscillator (VCO)  209 ′. A dual resonator VCO is required in order to switch between the two ensembles. A splitter  225 ′ supplies the output of the first mixer  208 ′ to first and second intermediate frequency (IF) amplifiers  227 ′ and  229 ′. The first IF amplifier  227 ′ is disposed in a terrestrial repeater signal processing path  231 ′ and the second IF amplifier  229 ′ is disposed in a second satellite signal processing path  233 ′. 
     In each path  212 ′ or  214 ′, a surface acoustic wave (SAW) filter is disposed. The first SAW filter  212 ′ isolates the signals from a selected ensemble received from a terrestrial repeater. The second SAW filter  214 ′ isolates the signals from a selected ensemble received from both satellites. The output of the first SAW filter  212 ′ and  214 ′ is input to a back end integrated circuit (IC) which mixes the filtered signal down from a first intermediate frequency (IF 1 ) to a second intermediate frequency (IF 2 ). For example, for the terrestrial arm  231 ′, IF 1  may be 209.760 MHz and IF 2  2.99 MHz. 
     In the satellite arm  233 ′, the SAW filter is adapted to isolate the signals from a selected ensemble received from both satellites. For the satellite arm  233 ′, IF 1  may be 206.655 MHz and IF 2  6.095 MHz. Those skilled in the art will appreciate that the present invention is not limited to the frequencies illustrated in the present disclosure. The outputs of the backend ICs  235 ′ and  237 ′ are output to analog-to-digital (A/D) converters as per the embodiment of FIG. 5 for digital processing. A channel decoder  300 ′ (not shown) digitally separates and decodes the two satellite channels. 
     In addition to the use of a single SAW filter to process the two satellite signals, a novel aspect of the embodiment of FIG. 6 is that since the satellite and terrestrial signals for ensemble A are the mirror image of the satellite and terrestrial signals for ensemble B, both signals can be received by using high side and low side injection into the first mixer  208 ′ using  221 ′ driven by the switched VCO  219 ′. See the above-referenced patent application Ser. No. 09/435,317 filed by P. Marko and A. Nguyen for a detailed discussion of this feature. 
     While the architecture of FIG. 6 is well adapted to receive a single ensemble at a time, in order to receive two ensembles at a time, it would be necessary to double the number of back ends (including the first mixer and every component thereafter). 
     FIG. 7 is a block diagram of a first embodiment of an SDARS receiver of the present invention. In the preferred embodiment, the full 12.5 MHz XM band containing the first and second ensembles are received in the receiver  200  via the antenna  110 , a low noise amplifier  122  and an image filter  124  as per FIG.  5 . The output of the image filter  124  is input to a first mixer  208 . The first mixer  208  is driven by a VCO  221  which, in the illustrative embodiment, operates at a frequency of approximately 1600 MHz. The actual output frequency of the VCO  221  will be substantially equivalent to two-thirds of the center frequency of the full 12.5 MHz frequency band received at the antenna  110 . If, for example, the center of the XM 12.5 MHz frequency band is 2338.750 MHz, the VCO should operate at two-thirds of 2338.750 MHz or 1559.167 MHz. The VCO is driven by a synthesizer  219 . 
     The mixer will have an approximate 800 MHz output which, in the illustrative embodiment, is filtered by a 12.5 MHz wide SAW filter  212 . Note that the use of a single SAW filter in place of the two SAW filters  212 ′ and  214 ′ of FIG. 6 is one advantage of the implementation of FIG.  7 . The SAW filter  212  serves to select the entire XM band  40  (see FIG. 3 a ) including both ensemble A and ensemble B. 
     The output of the SAW filter  212  is input to an automatic gain controllable (AGC) amplifier  228 . The gain of amplifier  228  is controlled by signal amplitude control stages (not shown) contained in demodulator blocks  317 ,  318  and  319 . The output of the AGC amplifier  228  feeds quadrature mixers  230  and  232 . The quad mixers  230  and  232  are driven in-phase at the IF frequency of 800 MHz with injection in quadrature. The injection signal is derived from the 1600 MHz signal output by the VCO  221  via a divide by 2 quad generator  234 . Hence, the quad generator  234  serves as a quad local oscillator operating at 800 MHz. 
     Recall that the output of the SAW filter is centered at 800 MHz in the illustrative embodiment. Consequently, the effect of mixing the output of the SAW filter with an 800 MHz signal is to mix the full 12.5 MHz band centered at the 800 MHz IF output of the SAW filter down to baseband (centered at 0 MHz IF). A graphical representation of this baseband signal can be seen in FIG. 3 b.  The two frequency slots assigned to satellite  12  are now centered at approximately ±5.2925 MHz, the two slots assigned to satellite  14  are centered at approximately ±3.4525 MHz and the two slots assigned to the terrestrial repeaters are centered at approximately ±1.2625 MHz. 
     Returning to FIG. 7, the outputs of the quad mixers  230  and  232  are amplified by post-mixer amplifiers  236  and  238  and input to low pass filters  240  and  242 , respectively. The quadrature (complex) baseband signals will have a bandwidth from 0 to ±6.25 MHz. Hence, the low pass filters should be designed to have a rolloff at a frequency of approximately 6.25 MHz or higher. The low pass filters  240  and  242  may be implemented with simplicity as one or two stage resistive/capacitive (RC) filters. 
     The filtered I (in-phase) and Q (quadrature) signals, output by the filters  240  and  242 , are digitized by analog to digital converters (ADCs)  224  and  226 , respectively. In the illustrative embodiment, the ADCs must at a minimum be capable of digitizing signals in the frequency range of 0 to 6.25 MHz. Those skilled in the art will appreciate that the outputs of the ADCs  224  and  226  constitute a digital complex baseband signal representing both ensembles (A and B) of the XM band and are ready for post processing. This digital representation can be applied to any of a number of digital selectivity elements. 
     In FIG. 7, the channel decoder  300  is shown as having three branches  302 ,  304  and  306  for processing the signal from the terrestrial repeater  16 , satellite  14  and satellite  12 , respectively. Since channel decoder  300  in FIG. 7 contains only three branches, only a single ensemble (A or B) at a time may be decoded. As each branch is similar (the filter bandwidth for the terrestrial repeater is wider than the bandwidth for the satellite), only one is described below for brevity. Each branch includes a complex mixer  311  which may be implemented with two mixers  312  and  313  driven by a complex numerically controlled oscillator CNCO  314 . The CNCO  314  is programmed to a frequency at the center of the frequency slot containing the satellite or terrestrial signal the branch is intended to receive. If for example branch  306  is intended to receive ensemble A of satellite  12 , CNCO  314  would be tuned to approximately −5.29 MHz. With CNCO  314  tuned to −5.29 MHz and applied to complex mixer  311 , the output of complex mixer  311  will contain the frequency slot assigned to ensemble A of satellite  12  centered at 0 MHz. 
     System controller  500  (of FIG. 5) also serves to select ensemble A or ensemble B for further processing by tuning the CNCO  314  to negative frequencies for ensemble A and to positive frequencies for ensemble B. 
     The digital low pass filters  315  and  316  act as channel or selectivity filters that remove the components relating to the other frequency slots in the 12.5 MHz band and any other residue that manages to pass the SAW filter  212 . Hence, at this point, the signal for each branch for the selected ensemble (A or B) is isolated and ready for demodulation (signal extraction) by demodulators  317 ,  318 , and  319  prior to being applied to a combiner  328 . The combiner applies error correction decoding to each of the demodulator outputs and takes the best of the three signals for output. 
     As illustrated in at the transport layer  320  in FIG. 5, in the preferred embodiment, the combiner uses a conventional Viterbi decoder (not shown) on soft decision bits from the first and second satellites  12  and  14  as, in the preferred embodiment, these signals are convolutionally encoded. Next, the Viterbi decoded signals are input to a Reed-Solomon decoder. The Reed-Solomon simply checks the validity or integrity of each codeword and applies corrections to a small percentage of errors. The RS decoded composite satellite signal is then ready for combination with the terrestrial repeater signal. (Those skilled in the art will appreciate that Viterbi decoders and Reed-Solomon decoders are well known in the art.) 
     Returning to FIG. 7, the stream at the output of the combiner  328  represents the bitstream that is to be multiplexed in the manner described more fully below. Those skilled in the art will appreciate that the receiver of FIG. 7 could be used to receive signals in the other assigned 12.5 Mhz band (presently allocated to CD Radio) by simply tuning to the ‘CD’ band centered at 2326.25 MHz instead of the XM band centered at 2338.750 Mhz. This would satisfy an FCC requirement that radios sold by both companies be fully compatible across the entire 25 Mhz digital broadcast spectrum. The digital filters would have to have a wider passband and the demodulators would have be changed to accommodate the CD Radio frequency plan. In an interoperable receiver, these changes could be realized with programmable filters and demodulators or with separate filter and demodulator paths, as will be appreciated by those skilled in the art. 
     FIG. 8 is an alternative embodiment of the SDARS receiver of FIG.  7 . The embodiment  200 * of FIG. 8 is essentially identical to that of FIG. 7 with the exception of the addition of a second VCO  235 * and a second synthesizer  237 *. In the illustrative embodiment of FIG. 8, the second VCO operates at 400 MHz. The use of two synthesizers eliminates the requirement that the 1 st  LO=2/3 the RF frequency. This allows for a lower frequency 1 st  IF which is programmable. 
     FIG. 9 is a block diagram of second alternative embodiment of the SDARS receiver of the present invention. The embodiment of FIG. 9 is essentially the same as that of FIG. 7 with the exception that each channel of each ensemble is provided for separately. That is, instead of simply retuning each CNCO from one ensemble to the other, three additional branches are provided  301 ″,  303 ″, and  305 ″ and each CNCO  314  is tuned to a different channel for a single ensemble. With additional demodulators  322 ″,  323 ″, and  324 ″ and an additional combiner  328 ″ the system is capable of receiving both ensembles simultaneously. Both ensembles are received simultaneously without replication of the front end circuitry including SAW filters, synthesizers and analog mixers. Another advantage of the architecture of FIG. 9 is that the signal processing is implemented in the preferred embodiment in digital complementary metal-oxide semiconductor (CMOS) technology. Those skilled in the art will appreciate that a significant advantage of a digital CMOS implementation resides in the fact that a digital CMOS implementation is on a very fast cost reduction path. 
     FIG. 10 is a block diagram of an alternative preferred embodiment of an SDARS receiver incorporating the teachings of the present invention. The receiver architecture  200 ′″ of FIG. 10 is similar to the receiver architecture  200 ″ of FIG. 9 with the exception that the receiver architecture  200 ′″ of FIG. 10 is a direct conversion architecture in which the SAW filter  212 ′ of FIG. 9 is eliminated. In addition, instead of using two local oscillators as per FIG. 9, the architecture of FIG. 10 employs a single local oscillator  221 ′″ which is driven to operate at twice the received frequency (e.g. 4800 Mhz in the illustrative embodiment) by a synthesizer  219 ′″ to provide a stable reference. (Those skilled in the art will appreciate that a crystal may be used for injection instead of a synthesizer, without departing from the scope of the present teachings, where the ability to move the reference frequency is not required.) The signal received by the antenna  110 ′″ is amplified by a low noise amplifier  122 ′″, input to a selectivity filter  124 ′″, amplified by an AGC amplifier  228 ′″ and applied to a quadrature mixers  230 ′″ and  232 ′″. Similar to the architecture of FIG. 9, the gain of amplifier  228  is controlled by signal amplitude control stages (not shown) contained in demodulator blocks  317 ,  318 ,  319 ,  322 ,  323  and  324 . 
     In the quadrature mixers  230 ′″ and  232 ′″, the RF signal, received at 2.4 GHz in the illustrative embodiment, is mixed with the 2.4 GHz quadrature local oscillator signals developed in quadrature generator  234 ′″ by dividing down the 4.8 GHz local oscillator signal. Consequently, the received RF signal is converted directly to baseband. With the direct conversion architecture of FIG. 10, no image filter is required (as would be the case with the superheterodyne receivers of FIGS. 7,  8  and  9 ) because the received signal is converted directly from RF frequency to baseband. 
     In each embodiment, the synthesizer outputs a reference frequency in response to the system controller  500  of FIG.  5  and thereby selects the XM radio band or the CD radio band of the digital broadcast spectrum as discussed above. 
     Returning to FIG. 10, the outputs of the quad mixers  230 ′″ and  232 ′″ are applied to post mixer amplifiers  236 ′″ and  238 ′″ and low pass filters  240 ′″ and  242 ′″. The low pass filters must be designed to handle the aliasing components which may be expected to result from an analog-to-digital conversion process implemented by ADCs  224 ′″ and  226 ′″. Low pass filters  240 ′″ and  242 ′″ will require a steeper rolloff than the low pass filters of FIG. 9, where additional anti-aliasing protection is available from SAW filter  212 ″. The output of the ADCs is a complex bit stream for processing in the manner described above with reference to FIGS. 8 and 9. 
     The architecture of FIG. 10 allows for the pursuit of improvements with respect to the tuner and the digital back end separately via a common interface  340 ′″. 
     Those skilled in the art appreciate that analog mixing of RF signals to complex baseband for digital conversion has inherent limitations related to the dynamic range of the input signals. In practice, these limitations often steer the receiver designer to digital conversion at an intermediate frequency, as described in the architecture of FIG. 6, at the expense of higher cost and size. One such limitation of mixing analog signals to baseband is second order intermodulation products generated in the baseband mixers and post mixer amplifiers. These undesired products develop when two RF (or IF) signal components (f 1  and f 2 ) present at the mixer input self mix and the difference product (f 1 -f 2 ) falls at baseband. If the amplitude of the difference product is sufficiently large, destructive interference with the desired baseband signal occurs. With the architecture of FIG. 7, SAW filter  212  protects the baseband mixers from strong interfering signals outside the XM band, which can create second order intermodulation products. Within the XM band, signals received from the satellites will have low signal amplitude which will not generate significant second order intermodulation products. In the scenario where the receiver is in close proximity to a terrestrial repeater, the repeater signal amplitude may be sufficient to generate significant second order intermodulation products. However, since the repeater signal contains program content identical to the satellite signal, in the event second order intermodulation products from the repeater interfere with the satellite signal, the signal recovered from the repeater will have more than sufficient amplitude to insure an error free bitstream is available to the end user. 
     With the architecture of FIG. 10, the SAW filter is eliminated and close-in selectivity for second order intermodulation protection from out of band signals is not available. However, by direct translation of the full XM frequency band to 0 Hz, the low amplitude satellite signals are isolated in frequency from most second order intermodulation products generated from out-of-band single carrier interferers, such as MCM carriers. This is evident by referring to the frequency plan of FIG. 3 b.  Since the satellite  14  and satellite  12  receive slots are centered at ±3.45 MHz and ±5.29 MHz, after digital translation the satellite signals may be separated from lower frequency intermodulation products with the digital complex mixers and low pass filters described previously. 
     A second limitation of analog mixing of RF signals to baseband is illustrated in FIG.  11 . FIG. 11 a,  two RF signals, S 1  and S 2 , centered at frequencies F 1  and F 2 , respectively, are depicted with S 2  having substantially larger amplitude than S 1 . Assuming S 1  and S 2  exist in the digital domain, FIG. 11 a  demonstrates the benefits of digital conversion to baseband. In FIG. 11 b,  a complex digital mixer has recentered the frequency band containing S 1  and S 2  to 0 MHz. Since digital mixers behave similar to ideal mixers, a substantially ideal replication of the RF spectrum exists at complex baseband after the digital frequency translation. 
     As depicted in FIG. 11 c,  the conversion of RF signals S 1  and S 2  to baseband using analog conversion results in the creation of images about 0 Hz axis due to gain and/or phase imbalance in the I and Q complex signal paths. The imbalance may be due to many causes including imperfect device matching, layout asymmetries, mechanical and process variations in present production RF circuit technology. Best case I/Q matching with standard bipolar integrated circuit processing results in a minimum image attenuation in the range of 30-40 dB. Referring back to the example depicted in FIG. 11 c,  the image of the large amplitude signal S 2  creates destructive interference for the small signal S 1 . Those skilled in the art appreciate that a receiver operating in a typical land mobile environment will encounter substantially large signal amplitude variations due to the varied proximity to terrestrial transmitters. A receiver architecture for multiple signal reception which includes an analog conversion to baseband stage would yield unacceptable interference protection due to the limited image rejection problem described above. The inventive receiver overcomes this limitation by symmetrically positioning the satellite signals about the 0 Hz axis. Since the XM satellite signals (or CD Radio satellite signals) are received on the ground with low margin (normally less than 15 dB), the signal dynamic range is limited such that the image created by a maximum amplitude satellite signal will not interfere with a low level satellite signal received at the minimum amplitude for detection. 
     FIG. 12 is a diagram showing an XM full waveform receiver adapted to receive audio and data simultaneously. The signal from antenna  110 ″ is received by the receiver  200 ′″ of FIG. 10 or the receiver  200 ″ of FIG.  9 . The outputs of the receiver  200 ′″ are first and second time-division multiplexed bitstreams A and B with approximately  100  channels of audio content and a number of data channels. The bitstreams are input to two types of demultiplexors broadcast  2010  and  2020  and data  2030  and  2040 . Through a switch  2050 , the user is able to select a broadcast channel from either ensemble A or B for listening pleasure as well as a data channel for informational purposes. 
     Returning briefly to FIG. 5, in the channel decoder IC the output of the combiner  328  is input to a service layer decoder  330 . In the service layer  330 , a demultiplexor  332  decrypts and extracts the desired channel information and provides digital audio and data to a separate source decoder  400 . The source decoder  400  provides digital audio to a digital-to-audio converter which applies an analog signal to an audio amplifier  840  and a speaker  860 . The data may be sent to a separate data interface  700  for external output or internal use. The system controller  500  has a man-machine interface  540  that controls the user interface  1000 . The interface  1000  also allows a user to control a conventional AM/FM radio, CD player or tape, the output of which is provided to the speaker  860  via the DAC  830  and amplifier/multiplexer  840 . 
     Thus, the present invention has been described herein with reference to a particular embodiment for a particular application. Those having ordinary skill in the art and access to the present teachings will recognize additional modifications, applications and embodiments within the scope thereof. 
     It is therefore intended by the appended claims to cover any and all such applications, modifications and embodiments within the scope of the present invention. 
     Accordingly,