Abstract:
Methods and systems are provided for reducing circuit area. Some embodiments provide electronic devices including an inductor formed from a path having two ends that loops substantially in a plane around a center area, wherein the path crosses itself at least two points and wherein the path defines an outer boundary of the inductor; and a circuit that is located within the outer boundary of the inductor and substantially within or adjacent to the plane. Other embodiments provide electronic devices including an inductor formed from a path having two ends that loops substantially in a plane around a center area, wherein the path defines an outer boundary of the inductor; and a circuit that is located within the outer boundary of the inductor and substantially within or adjacent to the plane, and wherein the circuit comprises a signal path that is rake-shaped and crosses the path of the inductor at substantially perpendicular angles.

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application is a continuation of International Patent Application No. PCT/US2006/020155, filed May 24, 2006, which claims the benefit under 35 U.S.C. §119(e) of U.S. Provisional Patent Application No. 60/684,496 filed May 24, 2005, each of which is hereby incorporated by reference herein in its entirety. 
    
    
     TECHNICAL FIELD 
     The disclosed subject matter relates to the field of integrated circuit design. More particularly, some embodiments of the disclosed subject matter relate to systems and methods for reducing circuit area. 
     BACKGROUND 
     An inductor is an electronic device used in circuits for its property of inductance. The behavior of inductors is related to phenomena associated with magnetic fields. When electric current flows through an inductor, magnetic fields may be created. Moreover, if the current varies with time, then the magnetic field will vary with time. This time-varying magnetic field induces a voltage in electrical conductors exposed to it. The circuit parameter of inductance relates the induced voltage to the current. 
     One property of an inductor is its quality factor, Q. The quality factor of an inductor is proportional to the ratio of its inductance to its resistance at a given frequency, and is a measure of the inductor&#39;s efficiency. The higher the quality factor, the more efficient the inductor is. The quality factor of an inductor may be influenced by several factors, one factor being eddy currents, which are the circulating flow of charges caused by a moving magnetic field within a nearby conductive material or device. The flow of eddy currents generates magnetic fields that oppose changes in external magnetic fields. Generation of eddy currents near an inductor degrades the inductor&#39;s quality factor, and thus if one is concerned with the quality factor of an inductor, it is usually desirable to avoid having eddy currents in devices near inductors. 
     On-chip inductors, despite being large, are often made of only two metal layers. A minimum metal density requirement for each metal layer over an entire wafer of an IC may be set to reduce topographical variations, increase uniformity, and target a certain yield. Therefore, it may be desired to increase the metal density count of a layer. Placing metal fills inside or near an inductor increases the metal density count, but has the possible drawback of decreasing the quality factor of the inductor through eddy current loss. Measurement results of inductors with metal fills have been reported before, but none have established a relationship between fill cell size and the inductor quality factor. This relationship is of great interest because it allows one to estimate the largest device that can be placed inside an inductor. 
     It should be noted that the topmost metal layer in a fabrication process is often referred as the thick top metal. The thick top metal layer is often several times thicker than the other metal layers, and thus has a lower resistance. This is beneficial for making inductors because a lower resistance improves the quality factor of an inductor. 
     Spiral on-chip inductors are a type of inductor used in the design of, for example, radio frequency integrated circuits (RFICs). These spiral on-chip inductors often occupy more than half of the total chip area in RFICs. The region in and around an on-chip inductor is typically kept clear of active and passive devices to avoid the generation of eddy currents in the devices, which, as discussed, degrades the quality factor of the inductor. However, leaving the area near an inductor empty is a waste of space and increases chip size. This is a problem because reducing the area, and therefore the cost of circuits is a concern in circuit design. 
     Therefore, it is beneficial to reduce restrictions on the spacing among devices on a circuit. Specifically, it would be of benefit to reduce spacing restrictions between inductors and other devices. 
     SUMMARY 
     Methods and systems are provided for reducing circuit area. Some embodiments provide electronic devices including an inductor formed from a path having two ends that loops substantially in a plane around a center area, wherein the path crosses itself at least two points and wherein the path defines an outer boundary of the inductor; and a circuit that is located within the outer boundary of the inductor and substantially within or adjacent to the plane. 
     Other embodiments provide electronic devices including an inductor formed from a path having two ends that loops substantially in a plane around a center area, wherein the path defines an outer boundary of the inductor; and a circuit that is located within the outer boundary of the inductor and substantially within or adjacent to the plane, and wherein the circuit comprises a signal path that is rake-shaped and crosses the path of the inductor at substantially perpendicular angles. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is an illustration of eddy current in metal fills inside a magnetic field. 
         FIG. 2  is a graph illustrating the relationship between inductor qualify factor and metal fill side length in accordance with some embodiments of the disclosed subject matter. 
         FIG. 3  is a photograph that includes two inductors that may be used in accordance with some embodiments of the disclosed subject matter. 
         FIG. 4  contains two graphs relating to eddy current loss with respect to the presence of metal fill. 
         FIG. 5  is a schematic diagram of a voltage controlled oscillator that may be used in accordance with some embodiments of the disclosed subject matter. 
         FIG. 6   a  is a simplified illustration of a layout style for a voltage tuning input of a voltage controlled oscillator and associated current flows in accordance some embodiments of the disclosed subject matter. 
         FIG. 6   b  is a simplified illustration of another layout style for a voltage tuning input of a voltage controlled oscillator and associated current flows in accordance with some embodiments of the disclosed subject matter. 
         FIG. 7   a  is a simplified illustration of rake-shaped metal wiring using for a voltage-tuning input in accordance with some embodiments of the disclosed subject matter. 
         FIG. 7   b  is a simplified illustration of a layout of a voltage controlled oscillator in accordance with some embodiments of the disclosed subject matter. 
         FIG. 8   a  is a graph illustrating the relationship between frequency and tune voltage in accordance with some embodiments of the disclosed subject matter. 
         FIG. 8   b  is a graph illustrating the relationship between output power and tune voltage in accordance with some embodiments of the disclosed subject matter. 
         FIG. 9   a  is a graph illustrating the relationship between phase noise and offset frequency in accordance with some embodiments of the disclosed subject matter. 
         FIG. 9   b  is another graph illustrating the relationship between phase noise and offset frequency in accordance with some embodiments of the disclosed subject matter. 
         FIG. 10  is a graph illustrating the relationship between phase noise and tune voltage in accordance with some embodiments of the disclosed subject matter. 
         FIG. 11  is a photograph of two voltage controlled oscillators, one with varactors and active devices outside the inductor and one with varactors and active devices inside the inductor in accordance with some embodiments of the disclosed subject matter. 
     
    
    
     DETAILED DESCRIPTION 
     One consideration in the design of integrated circuits is that, by keeping the sizes of devices in close proximity to an inductor small, the induced eddy current loops will be localized in small regions and thus the reduction of the inductor&#39;s quality factor will be lessened. A second consideration is that by carefully planning the current paths of devices, the magnetic coupling between the device currents and the inductor currents may be reduced. A further consideration is the relationship between the placement of metal fills in an around an inductor and the inductor&#39;s quality factor. 
     In accordance with some embodiments of the disclosed subject matter, systems and methods for placing passive and active devices inside an inductor are provided. In particular, non-active devices, such as varactors, as well as active devices are placed inside an inductor to create a compact voltage controlled oscillator (VCO) that has equal performance to traditional voltage controlled oscillators while using significantly less area. Some embodiments of the disclosed subject matter may also be applicable to other types of circuits. For example, the capacitors of a phase locked loop filter may be placed under a voltage controlled oscillator inductor. These systems and methods may result in the reduction of layout area and therefore the cost of circuits. 
       FIG. 1  illustrates eddy currents in metal fills inside a magnetic field. In this example, because the effect of metal fill structures with small dimensions is of particular interest, it may be assumed that the skin effect in the metal fill and the effect of the induced currents on the magnetic field can be neglected. 
     The circular metal fill  110  has a radius R 0    115 , and occupies about the same amount of area as N 2  smaller circular metal fills  120  with radii R 0 /N  125 . In this example, N has a value of 2. For mathematical convenience, circular metal fills are used to approximate square metal fills that are used in real layouts. Assuming a small fill cell size compared to the dimensions of the inductor, the magnetic field, B, is uniform over the area of the metal fill cell and is not a function of radius, r  135 . Radius, r  135 , is the radius that encloses the magnetic flux, (d), in a circular loop, where Φ is given by:
 
Φ= B·dA=B ·(π r   2 )  (I)
 
Using Faraday&#39;s Law,
 
                   V   =       -       ⅆ   Φ       ⅆ   t         =           ⅆ   B       ⅆ   t       ·   π     ⁢           ⁢     r   2                 (   II   )               
where V is the potential developed along any current path as a result of changing flux induced by AC current in the inductor. The negative sign indicates that the current, I eddy    140 , induced by V, will flow in such a direction as to oppose the flux that produced it. The resistance, R, of a thin cylindrical sheet of metal fill, bounded by the dashed and dotted lines  130  in  FIG. 1 , is equal to:
 
                     R   =       ρ     h   ·   dr       ·     (     2   ⁢           ⁢   π   ⁢           ⁢   r     )         ,           (   III   )               
where ρ is the resistivity of the fill metal, h is the height of the metal fill, and dr is the incremental thickness of the cylindrical sheet. From equation II and equation III, the total power dissipated in a metal fill with radius, R 0 , is:
 
                     P   diss     =       ∫       V   2     R       =     ∫           (       ⅆ   B       ⅆ   t       )     2     ·       π   ·   h       2   ·   ρ       ·     r   3       ⁢       ⅆ   r     .                   (   IV   )               
Since a large metal fill can be replaced by N 2  small metal fills, the power dissipation for the two cases are:
 
                     P   diss     =     {               (       ⅆ   B       ⅆ   t       )     2     ·       π   ·   h       8   ·   ρ       ·     R   0   4             (     in   ⁢             ⁢             ⁢   one   ⁢           ⁢   fill   ⁢           ⁢   with   ⁢           ⁢   radius   ⁢           ⁢     R   0       )                   (       ⅆ   B       ⅆ   t       )     2     ·       π   ·   h       8   ·   ρ       ·       R   0   4       N   2               (     in   ⁢           ⁢     N   2     ⁢           ⁢   small   ⁢           ⁢   fills   ⁢           ⁢   with   ⁢           ⁢   radii   ⁢           ⁢       R   0     N       )                     (   V   )               
The power dissipated in the metal fills is an additional loss mechanism for the inductor and thus reduces its quality factor. However, equation V shows, the power dissipation is reduced as the fill cell sizes are reduced. Accordingly, in some embodiments of the systems and methods of the disclosed subject matter, electric devices are divided into smaller electric devices to lessen adverse effects on other near-by devices.
 
     Equation V provides guidelines on device sizing inside an inductor, but is not sufficiently accurate for quantitative estimates. For more accurate estimates, full-wave simulations on inductors with different fill cell sizes may be run using an electromagnetic simulator such as ElectroMagnetic eXtractor (EMX) available from Integrand Software Inc. of Berkeley Heights, N.J. However, any suitable simulator may be used.  FIG. 2  illustrates the results from EMX simulations and shows a simulated quality factor at 2 GHz of a five-turn, 4.5 nH differential inductor, L diff , for different fill cell sizes and resistivity. Inductor L diff    510  is used in the VCO of  FIG. 5 , which will be described later. The inductor may be constructed using the thick top metal, and have an outer diameter of 200 μm, inner diameter of 80 μm, trace width of 10 μm, and trace spacing of 3 μm. The metal fills may be placed in the center of the inductor and may be constructed by stacking all the available metal and polysilicon layers without vias in between.  FIG. 2  illustrates the inductor quality factor versus square metal fill side length for multi-layer fills  210 , metal-only fills  220 , and polysilicon-only fills  230 . A metal fill side length of zero represents the case were no metal fills are used. The quality factor degrades rapidly as the metal fill dimensions become large, as can been seen only with multi-layer fills  210  and metal-only fills  220 . However, for the more resistive polysilicon fills  230 , the eddy loss is much less, as predicted by equation V. It should be noted that the term “M 6 ” refers to metal layer number six, and is the thick top metal layer of the fabrication process used in this embodiment. However, the number of metal layers can be altered without departing from the scope of the methods and systems of the present invention. It should also be noted that, in this embodiment, the multi-layer fills include metal layers M 1 -M 6  as well as a polysilicon layer. 
     The accuracy of EM simulations may be compared against measurement data from test structures, for example, a 0.25 μm BiCMOS process. A single-ended 2.3 nH inductor may be constructed using the thick top metal layer, with an outer diameter of 200 μm, and inner diameter of 100 μm, a trace width of 10 μm, and a trace spacing of 3 μm. A die photograph of such test structures is shown in  FIG. 3 . Inductor  310  is without metal fills, while the inductor  320  is with metal fills. In this example, the metal fills are 7 μm by 7 μm squares with spacing of 3 μm, and are of the multi-layer type. The metal fills are placed in area  340 , inside the inductor, and area  350 , around the inductor. 
       FIG. 4  illustrates quality factor versus frequency, in graph  410 , and the percent degradation in quality factor versus frequency, in graph  420 . The measured quality factor of plain inductor  310  is shown with the line formed of “o&#39;s”  430 . The measured quality factor of inductor  320  with metal fills is shown with the line formed of “*&#39;s”  440 . Lines  450  and  460  show the corresponding simulation results. An error of less than 5% in quality factor is observed between the simulated data,  450  and  460 , and the measured data,  430  and  440 , for a frequency below 3 GHz. A maximum quality factor degradation of approximately 10% occurs at its peak as shown in  FIG. 4  at reference label  470 . In contrast, the typical application range of an inductor in a tunable VCO is below its peak quality factor frequency since the varactors and the parasitics of the active devices add significantly to the tank capacitance. 
     The phase noise of voltage controlled oscillators (VCOs) is very sensitive to the quality factor of inductors. Because of this, VCOs are useful in demonstrating the systems and methods of the disclosed subject matter. Two identically designed VCOs having differed layouts may be used to demonstrate some of the advantages of placing a device inside an inductor in accordance with some embodiments. Specifically, VCOs may be placed inside an inductor for this purpose. As discussed above, however, placing transistors and varactors near the inductor will decreases the quality factor of the inductor due to the presence of eddy currents. However, by using a tank layout for VCOs in the inductor that places the transistors and varactors under the inductor as described above, significant losses due to eddy currents may be avoided. 
     Referring to  FIG. 5 , the layout of an inductor L diff    510  may be modified to allow placement of varactors  530  and  540  inside inductor  510 . As show in  FIG. 6 , by folding the leads of inductor  510  “outside in,” the cathodes of the varactors can be connected along the inner-most turn of the inductor. It should be noted that although this connection can result in a distributed effect, which is undesirable in a narrowband circuit, the actual effect on the circuit is small because the inner-most turn only contributes to a small fraction of the total inductance. The varactors  530  and  540  may be connected to the inductor  510  as illustrated in  FIG. 5 . It should be noted that the plurality of varactors  670  shown in  FIG. 6  are collectively shown as varactors  530  and  540  in  FIG. 5 . It should also be noted that the buffer  550  and the peak detector  560  are auxiliary circuits to facilitate measurements of the circuit. 
       FIGS. 6   a  and  6   b  illustrate two layout options for a V tune  line  520 , that connects the anodes of the varactors  530  and  540  together. In  FIG. 6   a  the anodes of the varactors  530  and  540  are connected together on the outside of the inductor. For example, at reference label  660 , the connection of the anodes of multiple varactors  670  is shown. A drawback of this configuration is that the current paths of V tune  line  520  are parallel to the flow of the inductor current, causing further unwanted magnetic coupling. The configuration illustrated in  FIG. 6   b  distributes V tune  line  520  from the center  650  of the inductor, thus keeping the wires with parallel current flow far apart. 
       FIG. 7   a  illustrates the details of rake-shaped metal wiring used to connect the anodes of the varactors  530  and  540 . The shape of the wiring is similar to that of a patterned ground shield (PGS). The fingers of the rake-shaped wiring are oriented such that the current flow is perpendicular to the direction of the inductor current, thus reducing magnetic coupling between the two. Furthermore, since the V tune  node is a signal ground for differential signals, the rake-shaped wiring acts as a grounded PGS that absorbs stray electric field from the inductor to the substrate, thus improving the quality factor of the inductor. 
     Another concern in the VCO tank layout is the resistance Of V tune  line  520 , since its series resistance adds thermal noise that is directly upconverted into phase noise. In order to address this, parasitic resistance extraction may be performed on the longest wire path from V tune  pin  520  to the varactor anode. In order to lower the wire resistance, multiple metal layers may be strapped together. Lowering the resistance of V tune  line  520  worsens its eddy current effect on the inductor, as Equation 5 suggests. However, the rake-shaped wiring in  FIG. 7   a  prevents eddy currents from circulating in large loops and thus reduces their effect. 
       FIG. 7   b  illustrates a layout of the VCO. Simulating the differential inductor, L diff    510 , by itself using EMX finds a quality factor of approximately 7 at 2 GHz. Small varactors of dimensions 12 μm by 11 μm, collectively shown as D 1   530  and D 2   540 , are connected to the differential inductor. In this case  82  varactor diodes are connected to the inductor, though any appropriate number may be used. The varactor diodes D 1   530  and D 2   540  may be made from p+ base in the n-well and have a simulated quality factor of about 40 at 2 GHz. It should be noted that the positions and connections of varactors  770  of  FIG. 7  substantially correspond to the positions and connections of varactors  670  in  FIG. 6B . The location of transistors  580  is also show in  FIG. 7   b.    
     Unlike the previously discussed metal fills, varactors actively participate in the circuit operation and carry AC currents. Their effect on the inductor is thus potentially more complicated than just extra loss due to eddy currents. For ease of simulation, the varactors may be replaced with parallel plate capacitors with similar plate resistivity as the varactor diodes while keeping the wiring the same. Simulations show that quality factor degradation is minimal when the varactors are placed under the inductor traces instead of at the center of the inductor. Not only are they exposed to a smaller magnetic field there, but they also perform the role of a PGS by further isolating the inductor from the lossy substrate. The active devices, which include a cross-coupled pair, current source, and current mirror may be replaced at the bottom center of the inductor. An EMX simulation may be run on the entire VCO structure which includes the differential inductor, the rake-shaped multi-layer metal routing, the parallel plate capacitors used to model the varactors, and the active devices. The extracted inductor S-parameters, which include all the eddy current effects, is used to evaluate the VCO in circuit simulations. Simulation results confirm that the performance of the compact VCO is close to the performance of the VCO with a plain inductor. 
     Benefits of embodiments of the systems and methods of the disclosed subject matter may be seen by constructing two VCOs, as discussed above, in accordance with the systems and methods of the disclosed subject matter. Referring to  FIG. 11 , the first is a VCO with varactors and active devices inside the inductor (hereinafter, “VCO IN”)  1110 . The second is a VCO with varactors and active devices outside the inductor, (hereinafter, “VCO OUT”)  1120 . Both VCO IN  1110  and VCO OUT  1120  may be implemented in a 0.25 μm BiCMOS process with only the peak detector implemented in bipolar transistors. Both VCOs, shown in the die photograph of  FIG. 11 , consume 3.2 mA from a 1.8V supply. The VCO output is connected to a buffer stage as well as a peak detector running off a 2.5V supply. Some embodiments may be implemented in any suitable transistor technology, including, for example, bipolar, CMOS, BiCMOS, and GaAs. 
     Properties of VCO IN  1110  and VCO OUT  1120  are illustrated in  FIGS. 8-10 . The information in these figures was generated by characterizing sets of VCO IN  1110  and VCO OUT  1120  with a Cascade RF probe station available from Cascade Microtech Inc. of Beaverton, Oreg. and an Agilent E4446A spectrum analyzer available from Agilent Technologies Inc. of Palo Alto, Calif. As shown, the measured data for VCO IN  1110  and VCO OUT  1120  was consistent despite the smaller area occupied by VCO IN  1110 . 
       FIG. 8   a  illustrates the tuning characteristics of VCO IN  1110  and VCO OUT  1120 . As illustrated, a wide turning range of 520 MHz, or 26% of the center frequency, was achieved.  FIG. 8   b  illustrates the output power for different tune voltages. Both VCO IN  1110  and VCO OUT  1120  have similar output power except for at high tune voltages, which corresponds to low output frequencies. 
       FIG. 9  illustrates the noise spectrums of VCO IN  1110  and VCO OUT  1120  for a 2 GHz carrier frequency and a corner frequency of approximately 300 kHz, as respectively shown by lines  910  and  920 . Line  930  has a slope of 1/f 2  and line  940  has slope of 1/f 3 . 
     Lastly,  FIG. 10  illustrates the variation in phase noise for different tuning voltages at various offset frequencies. The phase noise level at 600 kHz offset is shown for VCO IN  1110  and VCO OUT  1110  on lines  1020  and  1110  respectively. The phase noise level at 3 MHz offset is shown for VCO IN  1110  and VCO OUT  1120  on lines  1040  and  1130  respectively. As can be seen by reviewing  FIGS. 8 ,  9 , and  10 , VCO IN  1110  and VCO OUT  1120  have very similar performance. The die photograph of  FIG. 11  of VCO IN  1110  and VCO OUT  1120  illustrates that despite having the approximately the same performance, VCO OUT  1110  occupies 0.3×0.25 mm 2  while VCO IN  1120  only occupies 0.2×0.2 mm 2 . This size difference results in an area savings of 47%. As discussed, this area savings results in lowering the cost of manufacturing circuits. 
     Although the invention has been described and illustrated in the foregoing illustrative embodiments, it is understood that the present disclosure has been made only by way of example, and that numerous changes in the details of implementation of the invention can be made without departing from the spirit and scope of the invention. For example, although one example used was placing a VCO inside an inductor, the invention is not limited in this manner. Rather, according to various embodiments of the present invention, a phase locked loop filter or any other suitable circuit or combination of circuits may be placed under the VCO inductor. 
     Therefore, other embodiments, extensions, and modifications of the ideas presented above are comprehended and should be within the reach of one versed in the art upon reviewing the present disclosure. Accordingly, the scope of the present invention in its various aspects should not be limited by the examples presented above. The individual aspects of the present invention, and the entirety of the invention should be regarded so as to allow for such design modifications and future developments within the scope of the present disclosure. The present invention is limited only by the claims that follow.