Abstract:
Systems and methods of signal transmission and measuring for sensors employing a transmission medium are provided. In one embodiment, a method may comprise measuring a first monitored condition to generate a first monitored condition signal; converting the first monitored condition signal to a first frequency modulated signal having a first frequency; generating a second frequency modulated signal having a reference frequency; transmitting the first frequency modulated signal and the second frequency modulated signal using time division multiplexing; and wherein a first ratio of the first frequency and the reference frequency is associated with the first pressure.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation claiming priority under 35 U.S.C. §120 to U.S. patent application Ser. No. 13/252,840, filed Oct. 4, 2011, which is a continuation claiming priority to U.S. patent application Ser. No. 12/750,173, filed Mar. 30, 2010, now U.S. Pat. No. 8,122,770, issued Mar. 28, 2012, which is a continuation claiming priority to U.S. patent application Ser. No. 12/228,399, filed Aug. 11, 2008, now U.S. Pat. No. 7,685,880, issued Mar. 30, 2010, which is a continuation application claiming priority to U.S. patent application Ser. No. 11/803,128, filed May 11, 2007, now U.S. Pat. No. 7,409,866, issued Aug. 12, 2008, all of which are entitled “SIGNAL TRANSMISSION SYSTEM,” and all of which are incorporated by reference in their entirety as if fully set forth herein. 
    
    
     FIELD OF THE INVENTION 
     This invention relates to a signal transmission system for sensors and more particularly to a signal transmission and measuring system for sensors employing a single transmission wire and a grounded return. 
     BACKGROUND OF THE INVENTION 
     As one can ascertain, the prior art is replete with pressure transducers or sensors employed in harsh environments. Such environments include deleterious substances which may destroy the transducer, as well as high pressures and temperatures. High temperatures include those temperatures which are found in various high temperature environments as combustion engines, for example. In other applications, such as the use of pressure transducers in injection molding and for other environments extremely high temperatures are also found. The prior art has disclosed pressure transducers which are capable of operating at very high temperatures as temperatures in excess of six hundred degrees Celsius (600° C.). See for example, U.S. Pat. No. 7,124,639, which issued on Oct. 24, 2006, entitled “ULTRA HIGH TEMPERATURE HERMETICALLY PROTECTED WIREBONDED PIEZORESISTIVE TRANSDUCER,” by A. D. Kurtz et al. and assigned to Kulite Semiconductor Products, Inc., the assignee herein. See also U.S. Pat. No. 6,363,792, entitled “ULTRA HIGH TEMPERATURE TRANSDUCER STRUCTURE,” issued on Apr. 2, 2002 to A. D. Kurtz et al. and assigned to the assignee herein. See also U.S. Pat. No. 6,530,282, entitled “ULTRA HIGH TEMPERATURE TRANSDUCER STRUCTURE,” issued on Mar. 11, 2003 to A. D. Kurtz et al. and assigned to Kulite Semiconductor Products, Inc., the assignee herein. 
     By referring to the above noted patents, one can see applications of such transducers in high temperature environments as well as the monitoring of such signals in such environments. One problem is found when one deals in the oil and geothermal exploration fields. In such oil and geothermal explorations, one uses pressure or temperature transducers which are exposed to temperatures much higher than those experienced by standard electronics. Typical transducers which are used for normal operations are usually limited to temperatures below one hundred and twenty-five degrees Celsius (125° C.). Due to the depth of drilling as well as the use of steam to extract the oil the operating temperature in such explorations exceed two hundred degrees Celsius (200° C.). Pressure transducers using a piezoresistive silicon-on-insulator (SOI) structure are widely used in such applications. Such transducers for example are described in the above noted patents. Also used are platinum resistors (RTD) used to measure the temperature which resistors are also capable of operating at these high temperatures. Thus, the combination afforded in regard to the above is that one requires a pressure transducer which can operate at high temperatures and one also requires electronics which can operate at such temperatures. See for example a co-pending application entitled “HIGH TEMPERATURE PRESSURE SENSING SYSTEM,” U.S. patent application Ser. No. 11/234,724, filed on Sep. 23, 2005 for A. D. Kurtz et al. and is assigned to the assignee herein. In that application, there is described a high temperature pressure sensing system which includes a transducer having pressure sensing piezoresistive elements formed by a SOI process. The system also uses SOI CMOS electronic circuitry which is operatively coupled to the piezoresistive sensor as well as ancillary circuitry connected to the unit to provide compensation and normalization. That application is incorporated by reference in its entirety herein. 
     Other examples of SOI technology may be seen in U.S. Pat. No. 5,955,771, entitled “SENSOR FOR USE IN HIGH VIBRATIONAL APPLICATIONS AND METHODS OF FABRICATING THE SAME,” issued to A. D. Kurtz and U.S. Pat. No. 4,672,354. 
     In existing oil and geothermal applications, due to the depth of the drilling as well as due to the use of steam to extract the oil, very high temperatures are involved. In oil and geothermal explorations the wires used in these systems are extremely long and can be as long as ten thousand (10,000) meters. These wires apart from being extremely long are also expensive. The cost of the wire often exceeds the cost of the transducers. In prior art applications, the pressure transducers are connected to the wiring via a four-to-twenty milliamp electronic interface. The second wire is the metal conduit in which the wire is inserted. The prior art method has significant temperature limitations which are further aggravated by the significant power dissipation of the four-to-twenty milliamp interface. This power dissipation increases the junction temperature of the electronics by several tens of degrees above ambient temperature. The prior art method also requires a separate wire for each pressure or temperature sensor. 
     The present invention discloses a way of interfacing one or more pressure sensors to a measuring system using only one wire for the signal and power and a return wire which is usually the conduit of the signal/power wire. An electronic interface is advantageous for sensors located in a very high temperature environment at great distances from the measuring system such as described above in the oil and geothermal explorations. The invention is also well suited for integration in a circuit using technology suitable for high temperature operation as the silicon-on-insulator (SOI) process. The signal transmission system or wire interface described is also applicable and advantageous for use in systems operating at more benign temperatures and over shorter distances as it simplifies the wiring as well as the measuring method. 
     SUMMARY OF THE INVENTION 
     Apparatus for transmitting a transducer signal to be measured from a signal generation location to a measuring location connected by a single wire where undesirably the transducer signal is subjected to variations caused by multiple sources. The apparatus comprises a transducer positioned at the signal generation location and operative when biased by a power source to provide an output signal according to a monitored condition. A reference level generator is coupled to the power source and operative to provide a reference level output proportional to the value of the power source. A multiplexer for receiving at one multiplex input the transducer output signal and at another input the reference level output to provide at a multiplexer output the transducer signal for a first interval and the reference level for a second interval. A converter responsive to the multiplexer output for converting the transducer signal to a first frequency modulated signal having a frequency output variation according to the value of the transducer output signal during the first interval and for providing a second frequency modulated signal indicative of the reference level during the second interval, where any variations in signals which may be caused by multiple sources are present in both signals; and measuring arrangement positioned at the measuring location and responsive to the modulated signals to provide the ratio of the periods of the signals, where the ratio is a direct measure of the transducer output signal with the undesired variations substantially eliminated. 
    
    
     
       BRIEF DESCRIPTION OF THE FIGURES 
         FIG. 1  is a block diagram of a one wire system directed from a signal generation located to a measuring location according to an embodiment of the invention. 
         FIG. 2  is a block diagram of a wire interface located at the signal generation location according to an embodiment of the invention. 
         FIG. 3  consists of  FIGS. 3A and 3B  and depict timing diagrams as showing the pressure and reference intervals according to an embodiment of the invention. 
         FIG. 4  consists of timing diagrams showing the output of a capacitor and a monostable multivibrator operating according to an embodiment of the invention. 
         FIG. 5  is a block diagram of a signal measuring arrangement according to an embodiment of the invention. 
         FIG. 6  is an alternate embodiment of a signal measuring system according to an embodiment of the invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring to  FIG. 1  there is shown a block diagram of a one wire system for measuring a transducer output according to this invention. As part of the signal generation location there is a transducer system  10  which includes a bridge  11  and associated circuitry  12 . The bridge  11  is a Wheatstone bridge, which basically is implemented and fabricated by the use of piezoresistive SOI pressure transducers. The bridge  11  and associated circuitry  12  are located at the signal generation location. This location may be the bottom of a drilled shaft for oil exploration or for other purposes. Pressure sensors suitable for use in bridge  11  are well known, for example may be those and are described in the above noted patents and applications. The output of the bridge or pressure transducer can be compensated using its inherent resistance versus temperature characteristics to provide a stable ratio metric output over a wide temperature range. Transducers which operate accordingly are also well known in the art and are described for example also in the above noted patents. Coupled to the output of the transducer is an electronic interface  12 . The electronic interface may be fabricated and implemented by SOI electronic circuits including CMOS transistors. 
     The entire unit  10  as shown in  FIG. 1  which consists of the bridge  11  as well as electronics  12  may be positioned or inserted into a drilled shaft which would be implemented by techniques as for example used in oil well exploration. This shaft for example may be thousands of meters long. In any event, the entire signal generation apparatus  10  as shown enclosed in the dashed box is positioned near the bottom end of the shaft. The signal generation apparatus has an output  17  which is coupled to a wire  15 . The wire  15  runs from the transducer assembly  10  and is coupled to output  17  to a measuring site or location including a measuring circuit  18 . The length of the wire  15  may be ten thousand (10,000) meters or longer. Also shown is a return conduit  16  which may be a shield for wire  15  or may be an actual metal or other conduit used to surround and protect the wire. As seen at the measuring location which is the other end of the shaft and can be a field office or other ground location at normal ambient temperature is the measuring circuit  18 . The wire  15  is connected to a voltage source VMS (voltage at measuring source) via a resistor  19 . In the exemplary configuration, shown in  FIG. 1  VMS is five (5) volts while resistor  10  is one hundred (100) ohms. The values are by way of example only and other values can be employed. The VMS source supplies operating potential (+Vcc) to the transducer system  10  at the signal generation location. This VMS source is the sole power source used to bias the bridge  11  as well as to operate the circuitry  12 . The resistor  19  is DC connected to resistor  39  ( FIG. 2 ) associated with MOSFET  38  ( FIG. 2 ). In  FIG. 1  resistor  39  is shown, by way of example to be nine hundred (900) ohms. Thus, as seen in  FIG. 1 , the structure  10  which consists of the bridge  11  and the electronics  12  is located in a high temperature environment such as that found at the bottom of a shaft or hole drilled for oil or geothermal exploration and to measure pressure. The output  17  of the system  10  is directed to a single wire  15  which also is associated with a return shield or conduit  16 . The wire  15  can be more than ten thousand (10,000) meters long and is directed to the monitoring or measuring station whereby the output on wire  15  is measured to develop a voltage or an indication at measuring circuit  18  indicative of the pressure or other monitored condition. While the above noted system shown in  FIG. 1  depicts the measurement of pressure, it is understood that other measurements can be made such as temperature, etc., utilizing the single wire interface as described. 
       FIG. 2  shows a block diagram of the one wire interface showing the circuit details utilized to implement the signal generation system  10  of  FIG. 1 . Essentially before proceeding with a detailed discussion of the interface, a brief description of the operation will be given. 
     Referring to  FIG. 1  the one wire interface transmits the transducer output as a frequency modulated signal at output  17 . This frequency modulated signal propagates over wire  15  which also provides the power supply to the sensor and interface. Also transmitted over the wire  15  is a reference signal which reference signal is processed through the same chain as the transducer output signal. The reference signal is also a frequency modulated signal. The two signals are multiplexed in a time division mode, for example one second of the transducer signal followed by one second of the reference signal. The frequencies of the two signals are measured by the ground equipment as measuring circuit  18  and then the ratio of the two periods which are the reciprocals of the two frequencies is calculated. The ratio of the signals is a direct measure of the transducer output, and eliminates all sources of errors. Such errors can be significant, due to the high operating temperatures and are caused by multiple sources, as noise source, RF interference and the like. Therefore, it becomes very difficult to compensate for such sources. As the transducer output and the reference signal are passed through the same chain, and are affected by the same errors, the ratio calculation eliminates all errors. It does not matter how these errors are derived or generated, as they will be present both on the transducer output signal as well as on the reference signal and therefore can be eliminated by the apparatus and methods depicted herein. 
     As one will understand, the technique and apparatus can be further enhanced such that multiple sensors can be processed through the same chain and the data sent as time multiplexed signals followed by the reference signal. In this way several sensors can be connected to the ground measuring system  18  via a single wire  15 . It is also understood that temperature sensors can also be used with this interface. A RTD element can be mounted in a bridge configuration using three fixed metal film resistors and the output of the bridge multiplexed and sent through the same chain as the reference signal. The reference signal should be very consistent and stable with temperature in order to enhance operational effectiveness. 
     Referring to  FIG. 2  the reference signal is derived from the power supply +Vcc by a resistive divider which is located on the SOI chip. This implementation has shown that the ratio of resistors on the chip is very stable and consistent from chip to chip. This ratio is determined by the geometric features of the resistors and stays constant even though the value of the individual resistors may change over temperature and from device to device. Thus the techniques and apparatus described herein, also eliminates the errors due to variations in the supply voltage. Such variations are possible and expected due to the resistance of extremely long wires. As the transducer output and the resistive divider output which is the reference signal are proportional to the supply voltage +Vcc, the ratio calculation eliminates the undesired error as well. As one will understand the electronic interface as for example the circuitry  12  of  FIG. 1  is integrated on a SOI chip and the operating temperature of the interface exceeds two hundred and fifty degrees Celsius (250° C.), for example. As one can also understand, by referring to the above noted co-pending application entitled “High Temperature Pressure Sensor System” the SOI circuitry depicted therein can be employed herein as well, as for example, FETs, counters, and the like. 
     Referring to  FIG. 2 , there is a shown a circuit diagram in block form of the electronic interface  10 . As seen the transducer  20  is arranged as a Wheatstone bridge configuration. The Wheatstone bridge includes four piezoresistors such as  21 , which are wired in a bridge configuration. The bridge has a ratiometric output which is compensated over the temperature range. As seen the bridge derives its biasing voltage from the voltage source +Vcc which is applied to the bridge via a span resistor  22 . Thus the bridge  20  produces a ratio metric output which is compensated over the entire temperature range of operation. Such bridge circuits including those having ratiometric outputs are well known in the prior art and examples of such bridge circuits employing piezoresistors are indicated in the above cited patents. As seen the output of the bridge is applied to the input terminals of input (IN 1 ) of a multiplexer  26 . Also shown is a resistive divider consisting of resistors  23 ,  24  and  25 . The resistors  23 ,  24  and  25  are in series with one terminal of resistor  25  coupled to reference potential or ground and one terminal of resistor  23  coupled to the biasing voltage source +Vcc. 
     It is noted that the biasing source +Vcc for the resistive divider is the same biasing source employed for the bridge. The junction between resistors  23  and  24  is applied to one input terminal of the multiplexer  26  (INΦ) while the junction between resistors  24  and  25  is applied to the other terminal of the multiplexer input (INΦ). It is also noted before proceeding further that the resistors  23 ,  24  and  25  are also designated as R1, R2 and R3. The resistors have been so designated as their values are used in the mathematics which are pertinent to the operation of the system. The output of the multiplexer  26  (OUT) is applied to inputs of an instrumentation amplifier  27 . The output of the instrumentation amplifier  27  is applied to the non-inverting input (+) of an operational amplifier  28 . The operational amplifier  28  has a feedback resistor  30  also designated as RF which is coupled to a gain control resistor  29  indicated as R gain. The feedback resistor  30  is connected and at one terminal to the output of operational amplifier  28  and at the other terminal to the inverting input (−) of the operational amplifier  28 . Resistor  29  is coupled between the inverting input (−) of the operational amplifier  28  and reference potential. The total gain of the amplifier arrangement consisting of instrument amplifier  27  and the operational amplifier  28  is set by the resistor  29 . It is of course understood that the instrumentation amplifier  27  which is also an operational amplifier has a fixed gain such as a gain of ten (10) for example. The gain of the operational amplifier is controlled by the value of resistor  29  which is well known. 
     The output of the amplifier designated as V X  is coupled to the negative terminal of comparator  31 . The positive input terminal of the comparator is driven or coupled to a capacitor  37 . The capacitor  37  is charged by a current source  36  which is positioned in series with the source electrode of the FET  35 , also designated as Q1. As seen the current source  36  is also coupled to the +Vcc supply. The FET  35  also has its source electrode coupled to the positive input of comparator  31  and of course coupled to the non-ground terminal of capacitor  37 . The output of the comparator  31  is coupled to the input of a monostable multivibrator  33  whose output is coupled to the gate electrode of the FET  35 . 
     The output of the monostable is also coupled to a four plus eleven-bit (4+11-bit) counter and logic circuit  34 . The output of the logic circuit  34  is coupled to the selects input (S) of the multiplexer  26 . The output of the counter  34  is also coupled to the gate electrode of the FET  38  having a source electrode coupled to a load resistor  39 . The resistor  39  is coupled to wire  15  which at the other end has one terminal of resistor  19  coupled at the measuring end. The other terminal of resistor  19  is coupled to the VMS source. The junction between resistor  19  ( FIG. 1 ) and resistor  39  is the +Vcc which is the biasing potential shown in  FIG. 2  and used to bias all circuitry as amplifier  27 ,  28 , comparator  31 , mono  33 , counter  34  and so on. 
     Operation of the circuit is as follows. The positive input of the comparator  31  as indicated is driven by the capacitor  37 . The capacitor  37  is charged by the current source  36  and discharged by the MOSFET transistor  35 . The output of the comparator when present triggers the monostable circuit  33  when the capacitor  37  voltage reaches the predetermined value designated as V X . As one can see the output of operational amplifier  28  is V X . When the value of capacitor  37  is charged to V X  the comparator  31  produces an output which triggers monostable circuit  33 . The time period of the monostable multivibrator  33  is chosen to be as short as possible but long enough to safely discharge the capacitor  37 . In one particular example, a time duration of one microsecond for the output of the monostable  33  is appropriate. The value of the capacitor  37  and of the current source  36  is chosen such that they can be easily implemented on a SOI chip. The prescaler size included in module  34  is chosen such that the output pulses will have a sufficiently low frequency to provide a useful signal after passing through the very high capacitance and resistance of the very long single wire  15  connection to the measuring circuit  18 . The output of the monostable  33  as seen is applied to a fifteen-bit (15-bit) counter  34  (4+11 bits). The counter controls the select pin of the multiplexer  26  and also controls the gate of the switch MOSFET transistor  38 . The first four bits of the counter  34  are used as the prescaler and the following eleven (11) bits are used as a counter/sequencer. 
     The monostable output pulses are first divided by 16 and then the output is applied to the eleven bit sequencer. Thus, as indicated the counter controls the select pin of the multiplexer  26  as well as the gate of the MOSFET transistor  38 . When MOSFET transistor  38  is turned on the load resistor  39  is inserted in the circuit increasing the current consumption of the circuit. The increase in the current consumption results in a voltage drop of about one half volt across the one hundred (100) ohm resistor shown in  FIG. 1  and coupled to the input of the measuring circuit  18 . The eleven bit counter  34  controls the select pin of the multiplexer such that for one thousand and twenty-four (1,024) periods or the first interval of the sequencer the transducer output is processed by the interface through the inputs IN- 1  of the multiplexer  26 . After this period or interval the reference voltage of fifty millivolts (50 mV), which is derived from the voltage divider consisting of resistors  23 ,  24  and  25  is processed through inputs INΦ for another pulse period of one thousand and twenty-four (1,024) pulses of the prescaler. This is the reference interval. The output of the prescaler drives the gate of the MOSFET transistor  38  generating through the on/off switching of the load resistor  39 , the square wave current pulses, which appear as the voltage pulses at the input of the measuring system. 
     Thus, as seen in  FIG. 3A , during the first interval A, one thousand and twenty-four (1,024) measurement pulses are generated producing a first frequency modulated signal indicative of the value of the transducer output signal. During the next or second interval B, nine hundred and sixty (960) reference pulses are generated producing a second frequency modulated signal indicative of the reference level output. Thereafter, for a third interval C equivalent to a sixty-four (64) pulse period. Transistor  38  is disabled and no pulses are provided. This sixty-four (64) pulse interval informs the measuring system that the next sequence of pulses one thousand and twenty-four (1,024) is the transducer sequence. After counting one thousand and twenty-four (1,024) pulses the nine hundred and sixty (960) reference pulse interval begins and so on. It is also understood that the sixty-four (64) pulse period could be positioned between the one thousand and twenty-four (1,024) interval and the nine hundred and sixty (960) interval and serve the same purpose. These intervals are shown in the timing diagram of  FIG. 3A , the sixty-four (64) pulses are intended for allowing the measuring system to discriminate between the transducer signal phase and the reference signal phase.  FIG. 3B  is an expanded time scale showing transducer pulses T P  and the frequency variation as well as the reference pulses T R . 
     The voltage across capacitor  37  and the monostable output are shown in  FIG. 4 . Assuming that the capacitor  37  is discharged and a transistor  35  is off, the capacitor  37  is then charged linearly by the current source  36 . The capacitor charges until it reaches the voltage V X . At this moment the comparator  31  output changes state triggering the monostable  33  for a short period (e.g., about one microsecond). During this time the capacitor  37  is fully discharged. Afterwards the monostable  33  turns off, the transistor  35  is turned off and the cycle repeats. It is understood that the time scale shown in  FIGS. 3 and 4  is distorted for clarity purposes. The duration of the monostable pulses and the discharge time of the capacitor are less than that of a fraction of one percent (1%) of the charging time.  FIG. 3B  shows the pulses as depicted in  FIG. 3A  expanded in time. The pulse edges are not very fast, and in fact are slowed significantly by the very high capacitance of the long wire. Thus, as seen the pressure pulse designated at T P  has a relatively slow rise time and fall time as does the reference pulses designed as T R .  FIG. 3B  as indicated shows an expanded version of the pressure transducer pulse values as well as the reference pulse values depicted in  FIG. 3A . 
     Referring again to  FIG. 4 , there is shown the time diagram of the capacitor  37  voltage in the top diagram and the output of the monostable in the bottom diagram. Thus, as seen when the capacitor voltage reaches V X  the monostable multivibrator triggers for a duration of one microsecond. After the monostable pulse the cycle repeats again as depicted in  FIG. 4 . 
     Additional pressure transducers can be employed and for any additional transducer the multiplexer  26  will need an additional set of inputs and the counter/sequencer circuit  34  is implemented to provide additional intervals for the second, third and fourth transducer. As can be seen by referring to  FIG. 3A  one can implement multiple cycles concerning a cycle A, A 1 , A 2  followed by a reference cycle B. In the exemplary configuration, the value of resistor  19  of one hundred (100) ohms at the input of the measuring circuit is arbitrarily chosen. In a preferred embodiment, the value would be equal to the characteristic impedance of the wire and the conduit. In this case, the bandwidth of the signal transmission is significantly higher resulting in a much shorter measurement cycle than shown allowing multiple transducer data to be sent in a shorter time. Also, the pulses shown in  FIGS. 3A and 3B  will have much faster transition times resulting in a better accuracy and noise immunity of the period measurement. Another enhancement of the interface can be the addition of a thermal electric cooler for the electronic chip. This chip has a very small size and consequently a very small thermal mass. Thus, a small thermal electric cooler could be positioned on the chip to maintain the chip temperature at safe low levels without significant power consumption. Such thermal electric coolers also designated as PELTIER coolers are well known and are employed in many electronic chips such as microprocessors for use in computers and this will not be described in further detail. 
     In order to more clearly understand the nature of the invention the following circuit analysis is hereby presented. 
     Referring again to  FIG. 2 , in conjunction with  FIGS. 3 and 4 , assuming the sensor is a piezoresistive bridge  20  compensated using traditional ways, the bridge output voltage V BR  can be written as:
 
 V   BR   =k*p*V   CC ,
 
     where k is the bridge sensitivity, p is the pressure, and V CC  is the bridge supply voltage. During the measuring phase, i.e. when IN 1  of the mulitplexer  26  is selected, the output of the amplifier  28  V X  can be expressed as;
 
 V   X   =G*V   BR ,
 
     where G is the amplifier gain. 
     The voltage u c  across the capacitor  37  is: 
     
       
         
           
             
               
                 u 
                 c 
               
               = 
               
                 
                   
                     I 
                     Q 
                   
                   * 
                   t 
                 
                 C 
               
             
             , 
           
         
       
     
     where I Q  is the capacitor charging current, t is the time and C is the capacitance  37 . I Q  is generated by the current source  36 . When u c  reaches the level V X  then the comparator  31  changes state, thus triggering the monostable circuit  33  which rapidly discharges the capacitor  37  through the transistor  35  (Q1). Neglecting the very short discharge time, the cycle time T of the capacitor  37  (C) can be calculated by substituting V X  instead of u c , resulting: 
     
       
         
           
             T 
             = 
             
               
                 
                   C 
                   * 
                   
                     V 
                     X 
                   
                 
                 
                   I 
                   Q 
                 
               
               . 
             
           
         
       
     
     Taking into account the prescaler  34  factor of sixteen (16), the period T P  of the output signal during the measuring phase is: 
     
       
         
           
             
               T 
               P 
             
             = 
             
               
                 
                   16 
                   * 
                   C 
                   * 
                   
                     V 
                     X 
                   
                 
                 
                   I 
                   Q 
                 
               
               . 
             
           
         
       
     
     By substituting the formulas for V X  and V BR  the period T P  becomes: 
     
       
         
           
             
               T 
               P 
             
             = 
             
               
                 
                   16 
                   * 
                   C 
                   * 
                   G 
                   * 
                   k 
                   * 
                   p 
                   * 
                   
                     V 
                     CC 
                   
                 
                 
                   I 
                   Q 
                 
               
               = 
               
                 k 
                 * 
                 p 
                 * 
                 
                   
                     
                       16 
                       * 
                       C 
                       * 
                       G 
                       * 
                       
                         V 
                         CC 
                       
                     
                     
                       I 
                       Q 
                     
                   
                   . 
                 
               
             
           
         
       
     
     Considering now the second phase, when the reference voltage V R  is selected by the multiplexer  26  we have: 
     
       
         
           
             
               V 
               R 
             
             = 
             
               
                 
                   
                     R 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     2 
                   
                   
                     
                       R 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       1 
                     
                     + 
                     
                       R 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       2 
                     
                     + 
                     
                       R 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       3 
                     
                   
                 
                 * 
                 
                   V 
                   CC 
                 
               
               = 
               
                 r 
                 * 
                 
                   
                     V 
                     CC 
                   
                   . 
                 
               
             
           
         
       
     
     The factor r is the resistance ratio: 
     
       
         
           
             r 
             = 
             
               
                 
                   R 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   2 
                 
                 
                   
                     R 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     1 
                   
                   + 
                   
                     R 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     2 
                   
                   + 
                   
                     R 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     3 
                   
                 
               
               . 
             
           
         
       
     
     A similar calculation for the period T R  during this reference phase results: 
     
       
         
           
             
               T 
               R 
             
             = 
             
               r 
               * 
               
                 
                   
                     16 
                     * 
                     C 
                     * 
                     G 
                     * 
                     
                       V 
                       CC 
                     
                   
                   
                     I 
                     Q 
                   
                 
                 . 
               
             
           
         
       
     
     At this point, it is important to note that the capacitance C  37 , the gain G, the supply voltage V CC  and the charging current I Q  all have large random shifts with temperature and variations from device to device, while the factor r is very constant over temperature and from device to device. Calculating the ratio between T P  and T R  results: 
     
       
         
           
             
               
                 T 
                 P 
               
               
                 T 
                 R 
               
             
             = 
             
               
                 k 
                 r 
               
               * 
               
                 p 
                 . 
               
             
           
         
       
     
     This allows one to calculate the pressure p as: 
     
       
         
           
             p 
             = 
             
               
                 r 
                 k 
               
               * 
               
                 
                   
                     T 
                     P 
                   
                   
                     T 
                     R 
                   
                 
                 . 
               
             
           
         
       
     
     This formula shows that the pressure p can be calculated from two time measurements of T P  and T R  and from two very stable and well defined constants k and r. 
     For the component values shown on the schematic: 
     I Q =1 μA, 
     C=10 pF, 
     V X =1V to 4V during the measuring phase, 
     V X =2.5V during the reference phase, 
     resulting in the charging times of the capacitor to be one hundred microseconds (100 μs) to four hundred microseconds (400 μs) during the measuring phase and two hundred and fifty microseconds (250 μs) during the reference phase. 
     Due to the four-bit (4-bit) prescaler, i.e. divide by sixteen (16), the respective periods in the transmitted waveform are sixteen (16) times longer, i.e. T P =1.6 ms to 6.4 ms during the measurement phase, and T R =3.2 ms nominal during the reference phase. The duration of the flat portion of the waveform is sixty-four (64) periods of the reference pulses, corresponding to 204.8 msec. 
     It is important to note that the same calculations can be done for other measurements other than the pressure p. To measure the temperature, a bridge with three fixed resistors and an RTD as the fourth arm can be used. 
     As one can understand the interface as described above has extreme advantages over the prior art. One main advantage is that there is a reduced number wires for multiple transducers resulting in greater cost savings, less complexity, and improved reliability. The electronic circuit is implemented in a single integrated chip as all the components shown in the above diagram of  FIG. 2  are known building blocks of SOI processing techniques. Many of the components as seen in the block diagram exhibit large changes with temperature when processing a single signal. These errors are cancelled when the ratio of the two periods is calculated. Such errors result, for example, from mismatch of gain resistors, changes in the value of the capacitor (e.g., capacitor  37 ) changes in the value of the current source  36 , and changes in other circuit components from device to device as well as the effective resistance of the long wire and the value of the supply voltage. 
     As indicated, the circuit can be implemented as a very small integrated circuit chip with very few external connections. For a single transducer, the chip requires only five pins, allowing use of a very small package. The functions of the external measuring circuit are simpler than other implementations as the measuring circuit has to measure only two time periods which can be done with a comparator and digital circuits with a much better accuracy and much less complexity than any other measurement. The circuit can operate from a five volt supply reducing the power dissipation compared with other circuits requiring much higher voltages. There is no need for voltage regulators or stable references as all of the pertinent features due to changes and so on are cancelled by performing a ratiometric indication. 
     It is well known based on the above, as to how the ratiometric calculation can be performed as there are many circuits well known in the art which are capable of providing division of, for example T P  divided by T R  as well as multiplication. All of this can be implemented by a microprocessor or conventional circuits which are widely available. As discussed herein, the ratiometric measurement according to an aspect of the present invention cancels out substantially all variations. 
     Referring to  FIG. 5  there is shown a comparator  52 , the positive input or non-inverting comparator  52  is directed to resistor  51  where one terminal of resistor  51  connected to the +5 volt supply which is equivalent to the VSM supply depicted in  FIG. 1 . Resistor  51  is equivalent to resistor  19  shown in  FIG. 1 . The positive input of comparator  52  is directed to wire  15  and hence to the output of the sensor interface as depicted in  FIG. 2 . The inverting input of comparator  52  is biased by connecting it to the common terminal of the voltage divider consisting of resistors  53  and  54 . The voltage divider consisting of resistors  53  and  54  supplies a voltage of 4.75 or 0.25 volts below the supply voltage. This level corresponds to the middle of the pulses generated by the sensor interface. The pulses generated by the sensor interface, are shown in  FIG. 3B . 
     The output of the comparator is connected to an input of a microcontroller or microprocessor  55  which includes a timer  56  controlled by a crystal  57 . The measuring system as indicated is located remotely from the sensor interface and as depicted in  FIG. 1  is represented as measurement system  18 . The measuring system is at the surface of the ground if the interface  10  is placed in a drilled aperture or a drilled well associated with an oil well, for example. In any event, the measuring circuit is at a normal temperature such as room temperature or ambient temperature. The measuring system determines the two periods in the pulses present in one complete cycle and then calculates the ratio of these periods. This measurement can be done in many ways, using typical time and frequency measurements. Two such implementations are depicted in the figures and are based on identifying the start and end of the cycle from the flat portion (for example no pulses, corresponding in duration to the sixty-four (64) pulse duration) of the waveform. This is shown in  FIG. 3A . The sixty-four (64) reference level contains no pulses while the period indicative of the pressure transducer output contains one thousand and twenty-four (1,024) measurement pulses, while the duration of the reference signal is nine hundred and sixty (960) reference pulses. As indicated above, the period of the flat portion of the waveform is 204.8 milliseconds this is much longer than the periods of the measurement pulses which can vary between 1.6 milliseconds to 6.4 milliseconds or the reference pulses approximately at a period of 3.2 milliseconds. 
     Still referring to  FIG. 5 , the signal from the interface which is the signal shown in  FIG. 3  is applied to the positive input of the comparator  52 , while the negative input of the comparator is biased below the supply voltage at a level corresponding to the middle of the pulses generated by the sensor interface. This can be shown in  FIG. 3B . The output of the comparator is applied to the interrupt input (INT) of a microcontroller or a microprocessor  55 . The microcontroller  55  time stamps each interrupt by reading the internal timer  56  as well as the microcontroller counts these interrupts. By calculating the difference between the two successive interrupts, the microcontroller  55  identifies first the flat portion of the waveform, as its duration is much longer than any pulse in the sequence. After the flat portion is identified, the microcontroller  55  counts one thousand and twenty-four (1,024) pulses, determines their total duration and divides the result by one thousand and twenty-four (1,024), thus determining the period of the pulse as corresponding to the sensor data. Immediately after these one thousand and twenty-four (1,024) pulses, the microcontroller counts the next nine hundred and sixty (960) pulses. It then determines their total duration and divides the result by nine hundred and sixty (960) thus determining the period of the reference pulses. 
     The ratio of the two periods is then calculated to determine the value of the quantity to be measured. This, for example, may be pressure in the case of utilizing a pressure transducer or may be temperature in the case of using a temperature transducer. Either the pressure or the temperature transducer is arranged in a bridge circuit, as shown, for example in  FIG. 1 . Therefore, one can utilize this technique to measure pressure, temperature or any other value which can be implemented as a voltage at the output of a bridge configuration. 
     Referring to  FIG. 6  there is shown another exemplary method of measuring the output of the sensor interface. Also seen in  FIG. 6  there is a resistor  60  which is also equivalent to resistor  19  of  FIG. 1 . The resistor  60  has one terminal coupled to the input of analog-to-digital converter  61  and the other terminal coupled to the +5 volt supply. The input of the analog-to-digital converter  61  is connected to wire  15  and thus connected to the sensor interface. The output of the analog-to-digital converter  61  is connected to the input of a microcontroller  62 . The analog-to-digital converter  61  digitizes the incoming waveform. The sampling rate of the analog-to-digital converter is configured to be about ten (10) times that of the fastest pulses in the sequence resulting in a sampling interval of one hundred and sixty microseconds (160 us). The flat portion of the waveform is easily identified as no major transitions occur through relatively long duration of 204.8 milliseconds. 
     Next the period of a complete cycle is determined as the time between two successive flat portions. The waveform is then digitized for one complete cycle and the result stored. The computer then generates data for a theoretical waveform, with the same structure as the real one. Thus, for example, the computer generates one thousand and twenty-four (1,024) measurement pulses, nine hundred and sixty (960) reference pulses and a flat portion corresponding to sixty-four (64) reference pulses. The periods of the measurement and the reference pulses are arbitrarily chosen. The cross correlation function of the two waveforms is then calculated while the two periods in the theoretical waveform are varied until the cross correlation function shows a very short maximum value as a peak. The respective period values in the theoretical waveform corresponding to this peak represent the actual measurement and the reference period. Thus, the reference period again is used to produce the ratio between the pressure level period and the reference period to produce an output indicative of pressure, while undesired variations are thereby cancelled. 
     The programming regarding the microcontroller shown in  FIG. 5  and  FIG. 6  may be understood by one skilled in the art as the steps for producing and implementing the measuring system are clearly described. It is understood that there are other techniques which can be employed to measure the time period of both the frequency modulated pressure transducer output and the frequency modulated reference signal level output. 
     It should be understood by one skilled in the art that there are many alterations, and variations of the above noted circuitry all of which are deemed to be encompassed in the spirit and scope of the claims appended hereto.