Abstract:
A differential current steering (CS) circuit uses feedback from the differential output nodes A and B to cause current steering devices (e.g., MOSFETs) to effectively exhibit an infinite output impedance when conducting. Therefore, the signal on the output nodes A or B does not significantly change the voltage at the common node, This is particularly useful when the differential output nodes are connected to differential output buses in a digital-to-analog converter. The circuit dynamically cancels, though feedback, the signal induced at the common node by the signal present at the “steered” output node. Therefore, the CS circuit effectively presents an infinite output impedance between the common node and the output nodes. In some cases, it may be desirable to not create a substantially infinite output impedance for the CS circuit but control the impedance to a predefined level to counter other distortions in the system.

Description:
FIELD OF THE INVENTION 
     This invention relates to current steering circuits that provide a predetermined current to one of two differential output terminals in response to complementary input signals, such as for use in a digital-to-analog converter (DAC). 
     BACKGROUND 
     Current steering circuits are the basic building blocks of certain types of digital-to-analog converters (DACs) used in a wide range of applications, such as digital radio transmit signal chains, test equipment stimulus synthesis, wire-line data transmission, and so on. In these applications, a critical DAC specification is the spurious free dynamic range (SFDR), which is directly determined by the amount of undesired distortion unavoidably added to the generated output signal. 
       FIGS. 1A and 1B  illustrate an example of a prior art current steering (CS) circuit  100 . A fixed current source  130  is selectively steered towards the output node A or output node B as directed by complementary control signals VCA and VCB. 
       FIG. 1A  shows a first state of the CS circuit  100 . A current steering device  110  couples a current source  130  to output node A when signal Von is applied to node  112  through switching element  118 , as directed by control signal VCA present at control terminal  111 . The current steering device  110  may be any type of suitable device. In some embodiments, the device  110  may be a transistor, including n and p-channel enhancement and depletion MOSFETs, JFETs, MESFETs, heterojunction devices, NPN or PNP bipolar transistors, etc. The switching element  118  may be any type of signal generator that, in response to control signal VCA, provides a suitable bias to the current steering device. At the same time, current steering device  120  decouples current source  130  from output node B when signal Voff is applied to node  122  through the switching element  128 , as directed by control signal VCB present at control terminal  121 . In this first state of the CS circuit  100 , steering device  110  functions in a conductive state while steering device  120  functions in an isolating state. 
     A second state of the CS circuit  100  is shown in  FIG. 1B . The current steering device  110  decouples current source  130  from output node A when signal Voff is applied to node  112  through switching element  118 , as directed by control signal VCA. At the same time, current steering device  120  couples current source  130  to output node B when signal Von is applied to node  122  through switching element  128  as directed by control signal VCB. In this second state of the CS circuit, steering device  110  functions in an isolating state while steering device  120  functions in a conductive state. 
     The relatively low output impedance of steering device  110  in its conductive state is represented by resistor  115  in  FIG. 1A . The relatively low output impedance of steering device  120  in its conductive state is illustrated as resistor  125  in  FIG. 1B . Both current steering devices ( 110  and  120 ) in their isolating state have sufficiently high output impedance such that it can be practically ignored for the purpose of this description. 
     Because of physical implementation constraints, a relatively substantial parasitic capacitance is present at common node  132  and is illustrated by capacitor  135 . 
     In the first state illustrated by  FIG. 1A , a signal present at output node A may change the voltage stored on capacitor  135  through resistor  115 . Similarly, in the second state illustrated by  FIG. 1B , a signal present at output node B may change the voltage stored on capacitor  135  through resistor  125 . Such may be the case where the CS circuit  100  is one of many CS circuits in a DAC applying currents to a differential output bus connected to output nodes A and B. 
     When the CS circuit  100  transitions from the first state to the second state, the desired steering of current Io from output node A to output node B is accompanied by the undesired transfer to B of charge stored on common node  132  through resistor  115  by the signal present at node A during the first state. Similarly, when the CS circuit  100  transitions from the second state to the first state, the desired steering of current Io from output node B to output node A is accompanied by the undesired transfer to A of charge stored on common node  132  through resistor  125  by the signal present at node B during the second state. This transfer of charge creates distortion, limiting the spurious free dynamic range (SFDR) of the DAC. 
       FIG. 2  illustrates a DAC  200  constructed from a plurality of current steering circuits CS 1 , CS 2  . . . CS(n−1), CSn coupled to differential output nodes A and B. In one embodiment, these CS circuits are scaled (i.e., current sources increase for each bit position according to 2 0 , 2 1 , 2 2 , 2 3 , etc.) and are selectively controlled by complementary pairs of control signals VCA 1 , VCB 1 , . . . VCAn, VCBn to transition between first and second states at such time and in such sequence as to produce a desired differential output signal at nodes A and B. The differential signal thus produced at the output nodes A and B is an analog equivalent of the incoming control signals. In other embodiments, some or all of the CS circuits may be equal weighted or non-binary weighted or any combination thereof. 
     Due to the relatively low output impedance (represented by resistors  115  and  125 ) of the current steering devices  110  and  120  ( FIGS. 1A and 1B ) in their conductive state, the desired output signal is accompanied by undesired signal-dependent charge transfers, resulting in output signal distortion on nodes A and B. 
     What is needed is a current steering circuit, such as for use in a DAC, that generates less distortion of the signal at its output nodes. 
     SUMMARY 
     A differential current steering (CS) circuit uses feedback to cause the steering devices (e.g., MOSFETs) to effectively exhibit an infinite output impedance (in an ideal embodiment) when conducting. Therefore, the signal on the output nodes A or B does not significantly change the potential at the common node. This is particularly useful when the differential output nodes are connected to differential output buses in a DAC. 
     In one embodiment, the signal at output node B is scaled as a feedback signal and used to modulate the conventional control voltage for the steering device in the “A” leg when the steering circuit is controlled to steer current to output node A. The signal at node B is effectively the inverse of the signal at node A. The invention uses the inverse variable signal at output node B to cancel (or at least attenuate) the undesirable influence of the signal at output node A on the common node. The scaling (k) of the feedback circuit is designed, selected, and/or adjusted to prevent the variable voltages on the output nodes A and B from significantly changing the voltage at the common node. The result is equivalent to a substantially infinite output impedance for the CS circuit in the first state. 
     Similarly, the signal at node A is scaled as a feedback signal and used to modulate the conventional control voltage for the steering device in the “B” leg when the steering circuit is controlled to steer the current to the node B. The invention uses the inverse variable signal at output node A to cancel (or at least attenuate) the undesirable influence of the signal at output node B on the common node. The result is equivalent to a substantially infinite output impedance for the CS circuit in the second state. 
     In some cases, it may be desirable to not create a substantially infinite equivalent output impedance for the CS circuit but control the impedance to a predefined level to counter other distortions in the system. Therefore, the impedance may be intentionally under compensated or over compensated for an overall improved SFDR of the system. 
     Other embodiments are described. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1A  illustrates a prior art current steering circuit outputting a first differential signal. 
         FIG. 1B  illustrates the prior art current steering circuit outputting a second differential signal. 
         FIG. 2  illustrates a prior art digital-to-analog converter (DAC) with a plurality of current steering circuits where each receives complementary control input signals and outputs an analog signal representative of the control signals. 
         FIG. 3  illustrates a current steering circuit in accordance with one embodiment of the invention. 
         FIG. 4  illustrates a current steering circuit in accordance with another embodiment of the invention. 
         FIG. 5  illustrates a DAC with a plurality of current steering circuits in accordance with one embodiment of the invention. 
     
    
    
     Elements that are the same or equivalent are labeled with the same numeral. 
     DETAILED DESCRIPTION 
     The invention is an improved current steering circuit. The invention dynamically modulates a current steering device (e.g., a MOSFET, a bipolar transistor, an analog amplifier, or other device) while in its conductive state by using feedback from the differential signals present at the circuit&#39;s output nodes A and B, as illustrated in  FIG. 3 . In this way, the signal on the output node A or B does not significantly influence the potential at the common node. Hence, there is less undesired signal-dependent charge transfer, resulting in less output signal distortion on nodes A and B. 
     In the embodiments of  FIGS. 3 and 4 , the feedback circuitry in the CS circuits  300  and  400  appropriately modulates the signal applied to a steering device (e.g., a PMOS transistor) in its conductive state such as to maintain the voltage at the common node  332  or  432  independent of the signals present at output nodes A and B. 
     The term Von shown in  FIG. 3  represents a signal that is necessary to place the current steering device  310  or  320  in a conductive state, given the potential range expected or allowed at the output nodes A and B. The term Voff shown in  FIG. 3  represents a signal that is necessary to place the current steering device  310  or  320  in an isolating (non-conductive) state, given the potential range expected or allowed at the output nodes A and B. 
     In  FIG. 3 , let us assume that control signal VCA causes the switching element  318  to place steering device  310  in a conductive state while control signal VCB causes the switching element  328  to place steering device  320  in an isolating state. Hence the CS circuit  300  is in a first state corresponding to the first state of the CS circuit  100  shown in  FIG. 1A , and we will assume a rising signal is applied to output node A by parallel coupled CS circuits configured as illustrated in the DAC of  FIG. 5 . As a result of the finite impedance of steering device  310 , the potential of the common node  332  (having a parasitic capacitance represented by capacitor  335 ) will normally rise in response (although this effect is countered by the present invention). Simultaneously, due to the differential properties of the signal present at output nodes A and B, the potential of node B is decreasing. 
     In  FIG. 3 , the decreasing signal present at node B is shown coupled, through an appropriately scaled feedback circuit  317 , to the steering device  310  control node  312 . Thus, it reduces the conventional signal at node  312  such as to cancel the effect the increasing signal present at output node A has upon the common node  332 . The scaling (k) (or transfer function) of the feedback circuit is designed, selected, and/or adjusted to prevent the variable voltage on the output node A from changing the voltage at the common node  332 . The result is equivalent to a substantially infinite output impedance for the CS circuit in the first state. 
     In a similar fashion, when the CS circuit  300  is switched to its second state, the control signal to the current steering device  320  is modulated by coupling the node A signal through an appropriately scaled feedback circuit  327  to node  322 . The varying signal present at output node A proportionally modifies the conventional signal at node  322  of steering device  320  thus mitigating the effect of the inverse varying signal present at output node B. The scaling (k) (or transfer function) of the feedback circuit is designed, selected, and/or adjusted to prevent the variable voltage on the output node B changing the voltage at the common node  332 . The voltage at the common node  332  is independent of the voltage levels at nodes A and B due to the feedback. The result is equivalent to a substantially infinite output impedance for the CS circuit  300  in the second state. 
     A possible implementation of the circuit shown in  FIG. 3  uses a voltage controlled signal source as the feedback circuit  317  having a transfer coefficient k310, controlled by the node B signal, superimposed upon signal Von and applied to node  312 , as directed by control signal VCA. Thus, in the first state, signal Von+k310*V(B) is applied to node  312  in order to compensate for the effect of the potential at output node A on the common node  332 . 
     Similarly, a voltage controlled signal source may be used as the feedback circuit  327  having a transfer coefficient k320, controlled by the node A signal, superimposed upon signal Von and applied to node  322 , as directed by the control signal VCB. As a consequence, in the second state, signal Von+k320*V(A) is applied to node  322  in order to compensate for the effect of the potential at output node B on the common node  332 . 
     By designing, selecting and/or adjusting the transfer coefficients k310 and k320 based respectively upon properties of current steering devices  310  and  320  and the expected differential output signal amplitude, the potential of common node  332  can be maintained to be independent of the signals present at the output nodes A and B. Thus, the CS circuit  300  will appear to exhibit substantially infinite output impedance. 
     In certain applications it may be advantageous to under compensate or over compensate the effect of the output signal upon the current steering circuit common mode potential. This results in a tweaking of the current steering devices equivalent output impedance to a predefined level, which directly translates into a defined level of distortions introduced by the current steering circuit. Such a distortion control mechanism can be beneficially employed to counterbalance other distortion sources present in a system, resulting in improved overall spurious free dynamic range (SFDR). Toward this goal, the feedback coefficients (i.e. k310 and k320) may be selected to obtain the desired equivalent output impedance for the current steering devices, which may be different from infinity. 
     As it is well known in the art, in some applications signals Von and/or Voff may be embedded within the control signals VCA and VCB. In such configurations, VCA and VCB may be scaled analog signals coupled to nodes  312  and  322  through a variety of circuits such as amplifiers. 
     In some embodiments, the switching elements may be digitally controlled circuits (e.g. switches, transmission gates, . . . etc.) which, as directed by digital complementary control signals VCA and VCB, couple predetermined signals Von and Voff to the current steering devices such as to place them in the desired conductive and respectively isolating states. Thus, the feedback signal may be coupled through the switching element and modulate the predetermined Von signal. 
     In other embodiments, the switching elements may be analog driven circuits (e.g. amplifiers, buffers, attenuators, passive networks of resistors/capacitors/inductors or combinations thereof, . . . etc.) which translate (e.g. scale) and couple the complementary control signals VCA and VCB to the current steering devices such as to place them in the desired conductive and respectively isolating states. Thus the feedback may be coupled to the control terminals and suitably modulate the complementary control signals VCA and VCB. In such a case, the feedback circuits may scale the VCA and VCB control signals prior to the control signals being applied to a conventional current steering element. Suitable level shifting circuitry is well known. The overall effect is the same as in  FIG. 3 . 
     Another embodiment of this invention is illustrated in  FIG. 4  by current steering circuit  400 . Steering devices  410  and  420  are implemented using p-channel MOS transistors characterized at current Io, generated by current source  430 , by transconductance parameters gm410 and gm420, respectively, and by output conductance parameters gds410 and gds420, where gds=id/vds. In such a case, the gate voltages (at node  412 ,  422 ) of the MOS transistors are adjusted by the feedback circuits as described below. 
     Switching elements  418  and  428  are constructed using PMOS and NMOS transistor switch pairs  481 ,  483  and  482 ,  484 , respectively. The feedback circuits are implemented by resistor dividers RB 1 , RB 2  and RA 1 , RA 2 , respectively, comprising resistor pairs  471 ,  473  and  472 ,  474 , respectively. 
     During the first state, a “high” VCA control signal present at control node  411  places switch  481  in an isolating state and switch  483  in a conductive state. Consequently signal S 412  is coupled to node  412  where:
 
 S 412 =V on+( V ( B )− V on)* kb=V on*(1 −kb )+ V ( B )* kb  
 
and
 
 kb=RB 2/( RB 1 +RB 2).
 
     During the same first state, the complementary “low” control signal VCB present at control node  421  places switch  482  in a conductive state and switch  484  in an isolating state. 
     Thus signal V OFF  is coupled to node  422 , and steering device  420  is placed in an isolating state. 
     The potential variation (if any) of the common node  432  potential due to a signal variation v(A) present at output node A in the first state of the CS circuit  400 , without the benefit of the present invention, is proportional to v(A)*gds410/gm410. At the same time, due to the differential property of the DAC output signal, the variation v(B) present at output node B is approximately equal to −v(A). Therefore, by selecting kb=RB 2 /(RB 1 +RB 2 ) substantially equal to gds410/gm410, the variation of the common node  432  potential due the signal present at output node A during the first state can be eliminated or at least significantly reduced. 
     During the second state, a “high” VCB control signal present at control node  421  places switch  482  in an isolating state and switch  484  in a conductive state. Consequently signal S 422  is coupled to node  422  where:
 
 S 422 =V on+( V ( A )− V on)* ka=V on*(1 −ka )+ V ( A )* ka  
 
and
 
 ka=RA 2/( RA 1 +RA 2).
 
     During the same second state, the complementary “low” control signal VCA present at control node  411  places switch  481  in a conductive state and switch  483  in an isolating state. Thus, signal V OFF  is coupled to node  412 , and steering device  410  is placed in an isolating state. 
     The potential variation (if any) of the common node  432  potential due to a signal variation v(B) present at output node B in the second state of CS circuit  400 , without the benefit of the present invention, is proportional to v(B)*gds420/gm420. At the same time, due to the differential property of the DAC output signal, the variation v(A) present at output node A is approximately equal to −v(B). Therefore, by selecting ka=RA 2 /(RA 1 +RA 2 ) substantially equal to gds420/gm420, the variation of the common node  432  potential due to the signal present at output node B during the second state can be eliminated or at least significantly reduced. 
     One skilled in the art will recognize that this is a first order derivation of the feedback coefficients required in order to attain the desired substantially infinite output impedance for the current steering devices. A practical implementation presents a number of well documented higher order effects (e.g., body effect, intrinsic device capacitances, parasitic capacitances and so on) which must also be taken into account for a more accurate impedance control. 
     In some applications it may be desired to establish for the current steering devices a predefined equivalent output impedance goal which may be different from infinity. This predefined impedance level controls the amount of distortions introduced by the current steering circuit which may be used to counterbalance other distortion mechanisms present in the system for an overall improved SFDR. Thus the values of the feedback coefficients ka and kb may be selected higher or lower than calculated above. 
     The value of resistors  471  through  474 , the size of transistor switches  481  through  484 , as well as the size of steering transistors  410  and  420 , are further determined by the desired current steering circuit operation parameters according to methods well established in the art. 
     If the current steering device  410  is a PMOS transistor, as in the example of  FIG. 4 , the current through the transistor has two components: a first current determined by Vgs and a second current determined by Vds. The second current will change with a change in Vds (i.e., a change in signal at output node A). Because the sum of the two currents is constant (Io), the first current will change in the opposite manner. If the gate potential remains constant, this will cause Vgs to change, thus changing the common node  432 . The invention, in this example, ideally changes the gate potential exactly as much as necessary to compensate for the expected change in Vgs. 
     The feedback ratios ka and kb may be determined as a function of the properties of current steering devices  410  and  420  at design time or (in order to account for manufacturing process variations) may be adjusted using well known trimming techniques at manufacturing time. For even better performance (with respect to variations in temperature, power supply voltage, output load conditions, output signal amplitude, and so on), these feedback ratios may be further dynamically adjusted at power up or/and may be made to continuously track the properties of the current steering devices. The ratios may also be adjusted by the user by selecting appropriate values of components (e.g., resistors) connected to pins of an IC package, or by other means. 
     As an example of a preferred, but not limiting, implementation, resistors  471  and  472  may be fixed value resistors, while resistors  473  and  474  may be made (using well known active circuits) to track the properties (such as transconductance and output conductance) of replicas of the current steering transistors  410  and  420 . 
       FIG. 5  illustrates a differential DAC  500  comprising a plurality of CS circuits CS 1  through CSn steering currents to their respective output terminals A 1 ,B 1  through An, Bn, which are summed on the A and B output buses. Each CS circuit may incorporate a feedback configuration similar to that shown in  FIGS. 3 and 4 . The feedback transfer functions described above may differ for each CS circuit due to specific parameters and operating conditions of current steering devices comprised within. Accordingly, the level of the A and B buses does not affect the common node of each CS circuit. The result is a very low distortion DAC, enabling it to have improved SFDR performance. 
     While particular embodiments of the present invention have been shown and described, it will be obvious to those skilled in the art that changes and modifications may be made without departing from this invention in its broader aspects. The appended claims are to encompass within their scope all such changes and modifications as fall within the true spirit and scope of this invention.