Abstract:
A high voltage inverter is provided which includes a plurality of k-level flying capacitor H bridge modules, k being greater than 2, each having a positive dc terminal, a negative dc terminal, and two ac terminals, a connecting unit for connecting said ac terminals of said plurality of k-level flying capacitor H bridge modules in series to form a cascading set of modules, and a dc source connected to an ac source and having a transformer, a rectifier rectifying an output voltage of said transformer, and a capacitor connected between the positive and negative dc terminals.

Description:
FIELD OF THE INVENTION  
       [0001]    The present invention generally relates to power inverters and more particularly to a cascaded flying capacitor modular high voltage inverter. 
       BACKGROUND OF THE INVENTION  
       [0002]    Conventional medium voltage power inverter designs typically produce compromised results. For example, medium voltage drives and power supplies have been synthesized using high voltage devices, but these typically have high switching losses which adversely affects the output waveform harmonic content. 
         [0003]    A conventional variable frequency medium voltage drive typically is employed to drive a medium voltage motor for the control of 5 MW to 75 MW loads or processes. The conventional medium voltage drive usually operates at medium voltages of 4.16 kV to 13.2 kV as operation at voltages substantially lower than 4.16 kV leads to excessive currents and power losses. While a conventional medium voltage drive can be manufactured using a standard high voltage power semiconductor, the selection and availability of standard high voltage power semiconductors are limited. In addition, conventional high voltage power semiconductors typically are severely lacking in switching speed, thus limiting the switching frequency. In addition, conventional medium voltage drives typically require specialized multi-level circuitry that tends to increase the dollar cost per kva of the drive. 
         [0004]    Conventional low voltage drives of 400V-690V typically cost significantly less per kva than medium voltage drives due to higher production volume, technology maturity and market pressures. However, these conventional low voltage drives are not by themselves suited for controlling 5 MW to 75 MW loads or processes. Low voltage devices have been used in a series configuration, however, voltage sharing and balance during switching events is typically difficult and waveform harmonic content remain high. Conventional diode clamped and flying capacitor three-level converters allow a doubling of the dc bus and output voltage for a given switching device, but this voltage level is still limited by the device ratings. While the output harmonics are still excessive. Cascaded H bridge two-level converters, each fed by isolated direct current (DC) sources, have proven to be a suitable choice when a wide range of output voltage with high harmonic fidelity is desired. However, the number of cells required to synthesize medium voltages is excessive, and this directly correlates to an increase in the number of secondary transformer windings required to generate the dc source. 
         [0005]    There is a need for a medium voltage inverter topology that is not reliant on series devices. Further, there is a need to increase the voltage and switching frequency capability of existing diode clamped and flying capacitor solutions. There is also a need to limit the number of cascaded cells in a cascaded two-level H bridge converter without impacting the voltage and switching frequency capability. 
         [0006]    Thus, there exists a need for hybrid topology that increases the output voltage capability of each cascaded cell, thereby limiting the total number of cells, the associated rectifiers and transformer windings required to produce the isolated dc voltages. 
       SUMMARY OF THE INVENTION  
       [0007]    Methods and systems consistent with the present invention provide a medium voltage inverter having a variable voltage, variable frequency, and medium voltage output. The inverter may be, for example, a drive or power supply. The inverter beneficially includes a reduced number of cascaded stages required to synthesize a given output voltage level, without increasing the total harmonic distortion in the output waveforms. The inverter includes a plurality of multiple level flying capacitor inverter modules connected in series at their alternating current (AC) terminals to produce a desired AC output. Each k-level flying capacitor module can generate waveforms with the same harmonic fidelity as k-1 series two level H-bridges. The flying capacitor converter enables expansion above three levels at the cell level unlike conventional diode clamped converters. The availability of multiple voltages at the cell level, in combination with the cascaded structure provides increased flexibility in designing the system, while reducing the total number of cells. Each cell can accept a higher input DC voltage compared to conventional two-level inverters, yet may use the same switches as conventional two-level inverters. If this input voltage is generated from a transformer winding through a rectifier, the number of stages may be correspondingly reduced. 
         [0008]    In an illustrative example, three k-level cascaded flying capacitor poles may be connected to form a three-phase active rectifier which connects to the transformer secondary winding. In another illustrative example, two two-level active rectifiers can be connected in series to achieve DC bus regulation. If either of these active rectifier arrangements are used, bidirectional power flow can be attained and the regenerative operation is achieved. 
         [0009]    Other apparatus, features and advantages of the present invention will be or will become apparent to one having skill in the art upon examination of the following figures and detailed description. It is intended that all such additional apparatus, features and advantages be included within this description, be within the scope of the present invention, and be protected by the accompanying drawings. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS  
         [0010]      FIG. 1  is a schematic diagram of a cascaded flying capacitor modular high voltage inverter consistent with the present invention; 
           [0011]      FIG. 2  is a schematic diagram of three-level flying capacitor H bridge modules which may be cascaded to produce the inverter of  FIG. 1 ; 
           [0012]      FIG. 3  depicts a waveform generated by a three-level flying capacitor H bridge module; 
           [0013]      FIG. 4  depicts carrier waveforms for pulse width modulation for a three-level flying capacitor H bridge module; 
           [0014]      FIG. 5  is a schematic diagram of four-level flying capacitor H bridge modules which may be cascaded to produce the inverter of  FIG. 1 ; 
           [0015]      FIG. 6  depicts a waveform generated by a four-level flying capacitor H bridge module; 
           [0016]      FIG. 7  depicts carrier waveforms for pulse width modulation for a four-level flying capacitor H bridge module; 
           [0017]      FIG. 8  depicts a waveform generated by three cascaded three-level flying capacitor H bridge modules to produce an inverter consistent with the present invention; 
           [0018]      FIG. 9  depicts carrier waveforms for pulse width modulation for three cascaded three-level flying capacitor H bridge modules to produce an inverter consistent with the present invention; 
           [0019]      FIG. 10  is a schematic diagram of a cascaded flying capacitor modular high voltage inverter having zig zag transformer secondaries employed for the isolated DC sources; 
           [0020]      FIG. 11  is a schematic diagram of a cascaded flying capacitor modular high voltage inverter having zig zag transformer secondaries and 12 pulse rectifiers employed for the isolated DC sources; 
           [0021]      FIG. 12  is a schematic diagram of an k-level three-phase active rectifiers which may be used instead of passive rectifiers in accordance with the present invention; and 
           [0022]      FIG. 13  is a schematic diagram of stacked two-level three-phase active rectifiers which may be used instead of passive rectifiers in accordance with the invention. 
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       [0023]    Reference will now be made in detail to an implementation in accordance with methods, systems, and articles of manufacture consistent with the present invention as illustrated in the accompanying drawings. The same reference numbers may be used throughout the drawings and the following description to refer to the same or like parts. 
         [0024]      FIG. 1  shows an illustrative cascaded flying capacitor multi-level inverter  100  consistent with the present invention. The illustrative inverter is a three-phase wye connected inverter, where each phase  102 ,  104 ,  106  consists of a plurality of multiple voltage level flying capacitor H bridge inverter modules (“FCHBIM” or “module”) connected in series. In the illustrative example, phase A  102  includes modules  114  and  116 , phase B includes modules  118  and  120 , and phase C includes modules  122  and  124 . 
         [0025]    A load  108 ,  110 ,  112  is connected at to each respective phase  102 ,  104 ,  106 . The illustrative loads  108 ,  110 ,  112  are each shown as wye connected, however one having skill in the art would appreciate that a load may be delta connected. Each FCHBIM receives input voltage from a respective isolated DC source  126 ,  128 ,  130 ,  132 ,  134 ,  136  as shown. The total phase voltage applied to the load  108 ,  110 ,  112  is the sum of the voltages generated by each individual FCHBIM. As the flying capacitor multilevel topology allows each FCHBIM to generate theoretically unlimited AC output voltages, the number of FCHBIMs and isolated DC sources required may be reduced by increasing the number of levels within each FCHBIM. 
         [0026]      FIG. 2  shows a set of three-level FCHBIMs connected in series so a high output voltage can be developed at the load  224 . The module  202  is illustrative of each FCHBIM and includes eight switching devices  1 QL 1 ,  1 QL 2 ,  1 QL 3 ,  1 QL 4 ,  1 QR 1 ,  1 QR 2 ,  1 QR 3 , and  1 QR 4 , each of which may be for example an insulated gate bipolar transistor (IGBT), insulated gate commutated thyristor (IGCT), gate turn off thyristor (GTO) devices with antiparallel diodes, and the like. The eight switching devices are split into a left pole (including  1 QL 1 ,  1 QL 2 ,  1 QL 3 , and  1 QL 4 ) and a right pole (including  1 QR 1 ,  1 QR 2 ,  1 QR 3 , and  1 QR 4 ), each pole containing four switching devices Each pole further contains a capacitor  1 CL 1 ,  1 CR 1  referred to herein as the flying capacitor. 
         [0027]    For each pole, a first pair of switches is connected between a positive DC bus  204  and the AC output  206  ( 1 QR 1  and  1 QR 2  for the right pole). A second pair of switches is connected between the AC output  206  and a negative DC bus  208  ( 1 QR 3  and  1 QR 4  for the right pole). A first electrical connection of the flying capacitor  1 CR 1  is made in between the switches in the first pair (between  1 QR 1  and  1 QR 2  for right pole), and a second electrical connection is made between the switches of the second pair (between  1 QR 3  and  1 QR 4  for right pole). An energy storage capacitor  1 Cdc 1  is connected between the positive terminal  204  and negative terminal  208  of the DC bus. A common AC source  210  feeds separate transformers  212  whose secondaries are rectified through three-phase diode bridge rectifiers and source the DC bus capacitors  214 . 
         [0028]      FIG. 3  shows illustrative waveforms  302  generated by each FCHBIM of  FIG. 2  at its AC terminals for a sinusoidal reference voltage  204 . 
         [0029]      FIG. 4  shows illustrative carrier waveforms for generating the gate signals for each of the switching devices of  FIG. 2 . The illustrative carrier waveforms are generated with pulse width modulation with shifting of the respective carrier waveform. In the illustrative example, within each pole: 
         [0030]    Device  1 QR 1  and  1 QR 4  are oppositely phased, i.e. when  1 QR 1  is on  1 QR 4  is off and vice versa. Both devices are held off for a small time interval to prevent cross conduction. The time interval may be, for example, 1 to 30 μs. 
         [0031]    Device  1 QR 2  and  1 QR 3  are oppositely phased, i.e. when  1 QR 2  is on  1 QR 3  is off and vice versa. Both devices are held off for a small interval of time to prevent cross conduction. The time interval may be, for example, 1 to 30 μs. 
         [0032]    The gate signals for  1 QR 1  and  1 QR 4  are generated by comparing the reference sinusoid Vref 304 , to a triangular carrier waveform VtRO. The gate signals for  1 QR 2  and  1 QR 3  are generated by comparing the reference sinusoid Vref  304 , to a triangular carrier waveform VrRI which is phase shifted from VtRO by 180 degrees. 
         [0033]    The gate signals for  1 QL 1  and  1 QL 4  are generated by comparing the negative of the reference sinusoid Vref  304 , to a triangular carrier waveform VtLO. The gate signals for  1 QL 2  and  1 QL 3  are generated by comparing the negative of the reference sinusoid Vref  304 , to a triangular carrier waveform VtLI which is phase shifted from VtLO by 180 degrees. Also, the triangular carrier VtLO is phase shifted from VtRO by 90 degrees. 
         [0034]    Phase shifting of the triangular carriers as described above minimizes the harmonic content of the output AC waveform as well as maintains charge balance on the flying capacitors  1 CL 1 ,  1 CR 1 . 
         [0035]      FIG. 5  depicts another embodiment consistent with the present invention in which a set of four-level FCHBIMs are connected in series so a high output voltage can be developed at a load  504 . The module  502  includes twelve switching devices  1 QL 1 ,  1 QL 2 ,  1 QL 3 ,  1 QL 4 ,  1 QL 5 ,  1 QL 6 ,  1 QR 1 ,  1 QR 2 ,  1 QR 4 ,  1 QR 5 ,  1 QR 6  which can be, for example, an insulated gate bipolar transistor (1GBT), insulated gate commutated thyristor (IGCT), gate turn off thyristor (GTO) devices with antiparallel diodes, and the like. The eight switching devices are split into a left pole ( 1 QL 1 ,  1 QL 2 ,  1 QL 4 ,  1 QL 5 ,  1 QL 6 ) and a right pole ( 1 QR 1 ,  1 QR 2 ,  1 QR 3 ,  1 QR 4 ,  1 QR 5 , and  1 QR 6 ), each pole containing six switching devices each. Each pole further contains two capacitors referred to as flying capacitors  1 CL 1 ,  1 CL 2 ,  1 CR 1 ,  1 CR 2 . For each pole, the first set of three switches (e.g.,  1 QR 1 ,  1 QR 2  and  1 QR 3  for the right pole) is connected in series between a positive DC bus  506  and an AC output  508 . The second set of three switches is connected between the AC output and a negative bus  510  (e.g.,  1 QR 4 ,  1 QR 5  and  1 QR 6  for the right pole). The first connection of the first flying capacitor (e.g.,  1 CR 1 ) is made in between the first and the second switches in the first set of three switches (e.g., between  1 QR 1  and  1 QR 2  for the right pole) and the second connection is made between the second and the third switches of the second set of three switches (e.g., between  1 QR 5  and  1 QR 6  for the right pole). The first connection of the second flying capacitor (e.g.,  1 CR 2 ) is made in between the second and the third switches in the first set of three switches (e.g., between  1 QR 2  and  1 QR 3  for the right pole), and the second connection is made between the first and second switches of the second set of three switches (e.g., between  1 QR 4  and  1 QR 5  for the right pole). An energy storage capacitor  1 Cdc 1  is connected between the positive and negative terminals of the DC bus. A common AC source  512  feeds separate transformers  514  whose secondaries are rectified through three-phase diode bridge rectifiers and source the dc bus capacitors  516 . 
         [0036]      FIG. 6  shows illustrative waveforms  602  generated by each FCHBIM of  FIG. 5  at its AC terminals for a sinusoidal reference voltage  604 . 
         [0037]      FIG. 7  shows illustrative carrier waveforms for generating the gate signals for each of the switching devices of  FIG. 5 . The illustrative carrier waveforms are generated with pulse width modulation with shifting of the respective carrier waveform. In the illustrative example, within each pole: 
         [0038]    Device  1 QR 1  and  1 QR 6  are oppositely phased, i.e. when  1 QR 1  is on  1 QR 6  is off and vice versa. Both devices are held off for a time interval to prevent cross conduction. The time interval may be, for example, 1-30 μs. 
         [0039]    Device  1 QR 2  and  1 QR 5  are oppositely phased, i.e. when  1 QR 2  is on  1 QR 5  is off and vice versa. Both devices are held off for a time interval to prevent cross conduction. The time interval may be, for example 1-30 μs. 
         [0040]    Device  1 QR 3  and  1 QR 4  are oppositely phased, i.e. when  1 QR 3  is on  1 QR 4  is off and vice versa. Both devices are held off for a time interval to prevent cross conduction. The time interval may be, for example, 1-30 μs. 
         [0041]    The gate signals for  1 QR 1  and  1 QR 6  are generated by comparing the reference sinusoid Vref 604  to a triangular carrier waveform VtRO. The gate signals for  1 QR 2  and  1 QR 5  are generated by comparing the reference sinusoid Vref 604  to a triangular carrier waveform VtRII which is phase shifted from  1 VtRO by 240 degrees. 
         [0042]    The gate signals for  1 QL 1  and  1 QL 6  are generated by comparing the negative of the reference sinusoid Vref 604  to a triangular carrier waveform VtLO. The gate signals for  1 QL 2  and  1 QL 5  are generated by comparing the negative of the reference sinusoid Vref 604  to a triangular carrier waveform VtLI which is phase shifted from VtLL by 120 degrees.  1 QL 3  and  1 QL 4  are generated by comparing the negative of the reference sinusoid Vref  604  to a triangular carrier waveform VtLII which is phase shifted from VtLO by 240 degrees. Also the triangular carrier VtLO is phase shifted from VtRO by 90 degrees. 
         [0043]    Phase shifting of the triangular carriers as described above minimizes the harmonic content of the output AC waveform as well as maintains charge balance on the flying capacitors. 
         [0044]    By adding two controlled switches and a flying capacitor to each pole of a FCHBIM, the AC voltage can be increased. The number of levels in the output AC waveform also increases, thus ensuring that the harmonic content remains low. Thus, this strategy can be extended to a k-level FCHBIM to meet the cell AC output voltage requirements while maintaining low harmonic distortion, which provides performance flexibility at the module level. In an illustrative example, harmonic performance is optimized by shifting the k-1 carriers generated for each pole from one another by 360/(k-1) degrees. 
         [0045]    The FCHBIM modules can be cascaded to further increase the voltage capability of the inverter.  FIG. 8  shows illustrative AC output waveforms  802  generated by three cascaded three-level FCHBIM for a sinusoidal reference voltage  804 . To maintain harmonic and dv/dt control, the carrier waveforms in each cascaded stack are phase shifted from one another by 90/n degrees, where n is the number of FCHBIMs used. Three illustrative carrier waveforms are shown in  FIG. 9  for the three cascaded three-level FCHBIM implementation. Within each FCHBIM, the carrier waveforms obey the phase shifts as discussed above. 
         [0046]    To increase the harmonic performance on the input of the inverter system, zig zag transformer secondaries can be employed to create the isolated DC sources as shown in the illustrative example of  FIG. 10 . If higher level FCHBIMs are used, the higher DC voltage can be generated using multi pulse front ends as shown in  FIG. 11 . 
         [0047]    If regenerative operation is desired, the zig zag transformer of  FIG. 10  and the diode rectifiers can be replaced by n-level three phase flying capacitor active rectifiers as shown in  FIG. 12 . Alternatively, this regenerative operation can be realized by stacking standard two level three phase active rectifiers as shown in  FIG. 13 . 
         [0048]    The foregoing description of an implementation of the invention has been presented for purposes of illustration and description. It is not exhaustive and does not limit the invention to the precise form disclosed. Modifications and variations are possible in light of the above teachings or may be acquired from practicing the invention. The scope of the invention is defined by the claims and their equivalents.