Abstract:
The present invention relates to an arrangement and a method for amplifying high frequency RF signals having amplitude and phase variations. The objective problem of the present invention is to provide an efficient power amplifier for amplifying high frequency RF signals. The object is achieved by introducing a Signal Conditioning Device (SCD) that converts baseband signals SI and SQ into tri-states signals STI, STQ. The tri-states signals STI, STQ are time multiplexed, which results in that a single power amplifier can be used to amplify the signal on the In-phase (I) channel and the signal on the Quadrature (Q) channel. Hence, the present invention does not require a power combiner.

Description:
This application is the US national phase of international application PCT/SE02/01694 filed in English on 19 Sep. 2002, which designated the US. PCT/SE02/01694 claims priority to SE Application No. 0103412-3 filed 15 Oct. 2001. The entire contents of these applications are incorporated herein by reference. 
   FIELD OF INVENTION 
   The present invention relates to a power amplifier. In particular, it relates to a high efficient Non Linear Power Amplifier (NLPA) for amplifying high frequency Radio Frequency (RF) signals with varying phase and amplitude. 
   BACKGROUND OF THE INVENTION 
   High efficient Radio Frequency (RF) power amplifiers are critical components in modern radio communications systems. With worldwide increasing use of radio communications means, the energy consumption of very large number of radio base stations deployed becomes significant, which not only increases the cost for radio network operators, but also causes environment problems. 
   During the past, numbers of highly efficient power amplifiers have been proposed in literature. In particular, the switching mode power amplifier, such as class D, class E, class S, etc., have received the increasing attention for their possible applications in both radio receivers and radio base stations due to their very high efficiency. However, these types of switching-mode power amplifiers are highly non-linear. Special circuit architecture, such as, LInear amplification using Non-linear Components (LINC) technique, are usually required for their use in highly linear RF power amplifier applications. Although recent ideas of applying digitally, sigma-delta, modulated signals directly to switching-mode power amplifiers are promising. The operating frequency of these switching mode power amplifiers is, however, so far limited and not ready to be used in current radio transmitters with GHz carrier wave frequencies. The problem is that the high efficiency of these power amplifiers cannot be maintained due to the upper frequency limit of available RF power transistors and their intrinsic Ohmic losses with the current semiconductor technology. 
   In U.S. Pat. No. 3,896,395 two NLPAs  104 , 106  are used to amplify constant envelope RF signals directly after the mixers  108 ,  110  for both the I and Q branches, as shown in  FIG. 1 . The I and Q baseband signals S I , S Q  are converted into a bi-state (+1, −1) pulse train using a delta coder  112 ,  114 . This will generate a constant envelope PSK modulated RF signal alter mixing with the Local Oscillator (LO)  116 . However, a high level RF power combiner  102  used in the approach shown in this document may cause power lass (e.g., 3 dB loss), which win significantly reduce the total power efficiency of the linear amplification by Sampling Technique (LIST) as described in the document. 
   A drawback with the power amplifier disclosed U.S. Pat. No. 3,896,395, is that the power combiner  102  causes power loss as described above and hence degrade the efficiency. 
   SUMMARY 
   An objective of the present invention is to provide a power efficient power amplifier device by using a high efficient non linear power amplifier for amplifying a high frequency RF signal with phase and amplitude variations. 
   The power amplifier device provided by the present invention, comprising a Signal Conditioning Device (SCD), comprising means for generating an n-state level I component signal S TI  from said I component signal S I , means for generating an n-state level Q component signal S TQ  from said Q component signal S Q , and means for interleaving the n-state level I component signal and the n-state level Q component signal into, a non-overlapping in time, I component signal S TI  and Q component signal S TQ , a multiplexing device  204 , comprising means for generating a signal S MIQ  with a high frequency carrier wave wherein the I and Q components are being time multiplexed; and a single Power Amplifier (PA)  206  comprising means for amplifying said signal S MIQ , makes it possible to amplify a high frequency RF signal with phase and amplitude variations in a power efficient way. 
   The method carried out by power amplifier device provided by the present invention comprising the steps of:
     generating an n-state level I component signal from an I component signal S I ,   generating an n-state level Q component signal from a Q component signal S Q ,   interleaving the n-state level I component signal and the n-state level Q component signal into, a non-overlapping in time, I component signal S TI  and Q component signal S TQ ;   generating a signal S MIQ  with a high frequency carrier wave wherein the I and Q components are being time multiplexed and   amplifying said signal S MIQ  using a single Power Amplifier PA  206 , makes it possible to amplify a high frequency RF signal with phase and amplitude variations in a power efficient way.   

   An advantage with the present invention is that heatsinks, required by the output transistors in the amplifier, is smaller and thereby cheaper and more reliable (thanks to less thermal problems) in a high efficiency non-linear amplifiers. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  shows a circuit diagram of an example of prior art. 
       FIG. 2  discloses a circuit diagram of an example embodiment in a general mode. 
       FIG. 3  depicts a scheme of a RF power amplifier correlated with digital modulation according to an example embodiment. 
       FIG. 4  shows a scheme of a Pulse-Width Modulated (PWM) signal according to an example embodiment. 
       FIG. 5  depicts a scheme of a power amplification device with two switches and two NLPAS according to an example embodiment. 
       FIG. 6  depicts a scheme of a five-state controlled RF power amplification device with one switch according to an example embodiment. 
       FIG. 7  depicts a conventional digital modulation scheme according to prior art. 
       FIG. 8  depicts a scheme of a RF power amplifier comprising a predistortion unit according to an example embodiment. 
       FIG. 9  discloses an circuit diagram of a possible implementation of the Signal Conditioning Device (SCD) according to an example embodiment. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   The present invention amplifies a high frequency signal in a power efficient manner by using a high efficient power amplifier, which generally is non-linear, and by eliminating a power combiner  102  which causes power loss. 
   A general mode of an example embodiment is shown in  FIG. 2 . An amplification device  200  comprises a Serial to Parallel (S/P) converter  212  connected to two Band Limit Filters (BLF)  210 ,  214 . Each of the BLFs  210 ,  214  is respectively connected to a Signal Conditioning Device (SCD)  202 . The SCD is further connected to a multiplexing device  204  and the multiplexing device is further connected to a high efficient Power Amplifier (PA)  206  that is connected to a Band Pass Filter (BPF)  208 . 
   The S/P Converter  212  converts a Base Band Data (BBD) signal into two parallel datastreams, into an In-phase (I) respective a Quadrature (Q) component Each of the Band Limit Filters (BLF)  210 ,  214  averages the short bursts of different phase carriers into a continuous signal of an appropriate amplitude and phase. The output from the respective BLF  210 ,  214  is thus a respective baseband signal, S I  and S Q . There are however, many alternative ways in which the baseband S I  and S Q  signals can be generated for single carrier and multicarrier systems. 
   In the present invention, the use of the RF power combiner  102 , used in the U.S. Pat. No. 3,896,395, is eliminated. By the introduction of the SCD  202  the baseband signals S I , S Q  are converted into interleaved tri-state (−1, 0, +1) (or another odd number of states) signals, as shown in  FIG. 4 . Only one Non-linear Power Amplifier (NLPA) in  FIG. 1  is active at any given time, so the combiner  102  can be replaced by the multiplexing device  204 , and thereby eliminating the power loss. Since S I  is only in the active state (+1,−1 or non null (0) state) when S Q  is in the null state (0) and visa versa, it is possible to replace the two NLPA&#39;s  104  and  106  in  FIG. 1  by a single power amplifier  206  as shown in  FIG. 2 . 
   Signal Conditioning Device (SCD) 
   The SCD  202  generates in-time, non-overlapping switching signals S TI , S TQ  with a e.g. three-, or another odd number referred to as n, level state output. 
   Multiplexing Device 
   The multiplexing device  204  uses the S TI  and S TQ  control signals to time multiplex different components of the local oscillator signal (normally the I and Q components) into one signal S MIQ  so that a single amplifier can be used for both of the components. This is used instead of having a separate amplifier for the I and for the Q channel. 
   The power amplifier  206  is a highly efficient power amplifier, which generally is non-linear, and it amplifies the signal S MIQ . The purpose of the BPF  208  is e.g. to only permit signal bands of interest to pass and cut off all quantization noises generated by the SCD. The output signals after the BPF will then be the desired bandwidth-limited signals with both amplitude and phase modulations. 
   In  FIG. 3 , a first embodiment of the present invention is shown. The multiplexing device is implemented by means of multiplier devices MI and MQ, a phase shifter  302  and a Local Oscillator (LO). The LO generates the carrier wave with the frequency ω c  and the phase shifter shifts the phase of the carrier wave 90 degrees. Here, the output signals S MIQ  from the devices MI and MQ, which are controlled by S TI  and S TQ  respectively, will not overlap in time with each other, since the two multiplier devices MI and MQ are never enabled at the same time. If S TI  is separated from zero then S TQ  is always zero and visa-versa. Thus, the RF signal S MIQ  from the devices MI and MQ, may be a high-power RF signal and a Pulse-Width Modulated (PWM) signal, as shown in  FIG. 4 . 
   Pulse-Width Modulated Signal 
     FIG. 4  shows the PWM signals S TI  (on the I-axis) and S TQ  (on the Q-axis). The S TI  and S TQ  PWM signals have the states of +1, 0, −1 (n=3).
     S TI =+1:   
   When the S TI  channel is in +1 state, then an RF carrier is enabled (multiplied by +1) and therefore has 0 degree phase, see the multiplier device in  FIG. 3 .
     S TI =1   

   When S TI  is in the state of −1 the RF carrier is multiplied by −1 and has therefore a phase shift of 180 degrees.
     S TI =0   

   When S TI  is in the 0 state then the RF carrier is multiplied by 0, i.e. nothing comes out from the multiplier.
     S TQ =+1   

   When the S TQ  channel is in +1 state, then a 90 degree phase shifted RF carrier is enabled (multiplied by +1) and therefore has 90 degree phase, see the multiplier device in  FIG. 3 .
     S TQ =−1   

   When S TQ  is in the state of −1 the 90 degree phase shifted RF carrier is multiplied by −1 and has therefore a phase shift of 90−180=−90 degrees.
     S TQ =0   

   When in the 0 state then the 90 degree phase shifted RF carrier is multiplied by 0 i.e. nothing comes out from the multiplier. 
   Thus, highly efficient nonlinear power amplifiers, such as class C, D, E or S, can be used to amplify these signals, since the signals have constant envelopes. The SCD  202  may use noise shaping techniques to improve performance by shifting quantisation noise away from the desired signal spectrum. An example of such a noise shaping technique is the “sigma delta” or “delta sigma” method used in analogue to digital converters. 
   In  FIG. 5 , a second embodiment of the present invention shows a possible approach of implementing the MI and MQ multiplier devices by using tri-state switches. Here, the switching frequency for respective device MI and MQ, controlled by respective signal, S TI  and S TQ , is in the order of several hundreds of MHz, depending on the bandwidth of the baseband signal which is usually on the order of a few tens of MHz. At these operating frequencies, semiconductor mixers and switches are readily available. It is also shown in  FIG. 5  that the amplification of the RF signals can be split between two different amplifiers PA 1 , PA 2 . This implies that the normal low-level power carrier wave, with the frequency ω c  generated by the LO first is amplified by using a high-efficient NLPA PA 1 . This is possible because the carrier wave ideally is an unmodulated sine wave that can be amplified by an NLPA without causing problems of generating intermodulation products. Depending on available power-handling capability of switching devices, their insertion loss, as well as other possible implementation issues, further power amplification PA 2  of the S MIQ  signal can be made just prior to the BPF as disclosed in  FIG. 5 . 
   Further, in an amplifier device  600  shown in  FIG. 6 , only one multi-throw switch MC is used to perform the required PWM of a, e.g. multiphase, signal. The number of throws in the switch is herein referred to as n. To control the switch, it is required by the SCD  602  to generate a five-state (n=5) control signal S c , (because of the five throw switch), as shown in a third embodiment of the present invention in  FIG. 6 . S c  is a five state control line (digital signal) that controls which RF phase shift (or null signal), that is sent to the power amplifier PA 2 . In order to generate the control signal S c , the two three-state outputs from the SCD, i.e. S TI  and S TQ  showed in the previous embodiments, may be passed into a signal translation unit (not shown in the drawings), with S c  as the output signal. The signal translation unit has the following function: 
   
     
       
             
             
             
             
           
             
             
             
             
           
         
             
                 
                 
             
             
                 
               S TI   
               S TQ   
               S C   
             
             
                 
                 
             
           
           
             
                 
             
           
        
         
             
                 
               0 
               0 
                0 degrees (MC is in the top position, see FIG. 6.) 
             
             
                 
               0 
               −1 
               270 degrees (MC is in the second position down.) 
             
             
                 
               −1 
               0 
               180 degrees (MC is in the third position down.) 
             
             
                 
               0 
               +1 
                90 degrees (MC is in the fourth position down.) 
             
             
                 
               1 
               0 
                0 degrees (MC is in the fifth position down, 
             
             
                 
                 
                 
               i.e. bottom position.) 
             
             
                 
                 
             
           
        
       
     
   
   Other combinations of S TI  and S TQ  are not possible because of the design of the SCD. It is also possible to use implementations with a greater number of amplitude and phase states, i.e n&gt;5. 
   It may be noticed that the RF signal S MIQ , shown in  FIG. 3 ,  FIG. 5  and  FIG. 6  is different from the RF signal S out  from the conventional digital modulators as shown in  FIG. 7 . In general, the conventional digital modulated RF signal S out  can be both amplitude and phase modulated with a time-varying envelope. For such a signal, it is not possible to use NLPA. This is due to undesired intermodulation products being generated by the NLPA which results in a great reduction of the quality of the output signals, which has to fulfil certain linearity requirements. 
   Considering potential practical implementation issues, a known predistortion technique may be introduced, as shown in  FIG. 8 . A predistorter unit  802  is connectable to the respective BLF  810 ,  814  and to the SCD  804 . The predistortor unit comprises two inputs for the I- and Q-signals S I , S Q  and one input for the amplified signal S showed in  FIG. 8 . I.e. the amplifier device comprises a feed-back loop from the amplified output signal to the predistortor unit  802 . It further comprises two outputs for the I- and Q-signals S I , S Q , wherein each output is connected to the SCD  804 . The predistortor unit  802  may comprise of lookup tables and other means for adjusting. The predistortor unit  802  compensates for any non-linearities and other errors that may distort the signal during the conversion and amplification process. This is to further improve the signal quality, depending on specification on the linearity of the output RF signal. 
   Furthermore, to improve the efficiency of the RF power amplifier, some energy recovery techniques can be applied to take care of some reflected reactive RF power from the circuits. 
   Signal Conditioning Device 
   In  FIG. 9 , an example of a possible implementation of the SCD  900  is disclosed. Sigma delta Σ-Δ encoders for the I and Q channels spectrally shape the noise away from the desired signal. The output levels of +1, 0, −1 are enabled by the three-state output from the sigma delta encoders. A controlling device is used on the output from the three level decision devices to stop simultaneous |1| outputs on the S TI  and S TQ  lines. The controlling device will only allow a non-zero signal on the output S TI  line if |x i | is larger than |x q | at the decision time set by the pulse enabled by the clock. The switch positions are shown in the position where x q  is larger that x i , allowing output from the Q channel three level decision device on the S TQ  line. The controlling device is embedded in the Σ-Δ feedback loops. 
   The classical digital modulator architecture is changed in such a way that RF power amplification can be made efficiently by use of highly efficient non-linear power amplifiers. Non-linear problems e.g., intermodulation products is eliminated or greatly reduced for complex digital modulated signals with both phase and amplitude modulations such as Wideband Code Division Multiple Access (WCDMA) and other multi-carrier signals. 
   The present invention is not limited to the above-described preferred embodiments. Various alternatives, modifications and equivalents may be used. Therefore, the above embodiments should not be taken as limiting the scope of the invention, which is defined by the appending claims.