Abstract:
A voltage level converter for converting an input signal at a first voltage level to an output signal at a second voltage level, the converter comprises: an input for receiving said input signal; an output for outputting said output signal; a circuit node; precharge means for charging or discharging said circuit node to a third voltage level during a first time period by connection of said circuit node to a first voltage supply; isolation means for isolating said circuit node from said first voltage supply during a second time period; input means for changing the voltage at said circuit node in dependence on the voltage at said input during a third time period; and output means arranged so that the voltage at said output depends on the voltage at said circuit node.

Description:
TECHNICAL FIELD OF THE INVENTION 
     The invention relates to voltage level converters, and is useful in inter alia high speed CMOS circuits which receive many data inputs at a voltage significantly less than the supply voltage. The invention is of particular use in the field of monolithic digital data drivers for active matrix displays. In these circuits low performance TFT circuitry requires a high supply voltage to achieve the necessary speed of operation, whereas to minimise power consumption low voltage inputs are desirable. 
     DESCRIPTION OF THE RELATED ART 
     The simplest. type of prior art digital level shifter is shown in FIG. 1, and described in detail in A. Bellaouar, M. Elmasry.  Low Power Digital VLSI Design . Kluwer Academic Publishers, 1995. The circuit consists of two CMOS inverters  2  and  4 , each comprising a p-type (PMOS) transistor M 1 , M 3  and an n-type (NMOS) transistor M 2 , M 4 . The first CMOS inverter  2  is powered by a supply voltage VDD 1 , and the second CMOS inverter  4  is powered by a larger supply voltage VDD 2 . The main disadvantage of this circuit is that the power consumption is considerable since both transistors M 3  and M 4  in the second inverter  4  conduct when the output of the first inverter  2  is high, ie. equal to VDD 1 . One of the basic criteria for minimising power consumption is that there should be no direct current path between the supply voltages and ground. 
     A further prior art circuit is shown in FIG. 2, and described in detail in U.S. Pat. No. 4,486,670. This circuit overcomes the problem of direct power consumption by using a latch (consisting of the four transistors M 5 , M 6 , M 7  and M 8 ) to ensure that if one of the complementary transistors M 3  or M 4  is turned on, the other is turned off. 
     A widely used level shift circuit with improved latching action is shown in the prior art circuit of FIG. 3, and described in detail in U.S. Pat. No. 4,845,381. Two CMOS inverters  6  and  8 , made up of transistors M 1 , M 2  and M 4 , M 5  respectively, receive complementary digital input signals INPUT and INPUT*, varying between voltages of VDD 1  and GND. The outputs  9  and  11  of the inverters  6  and  8  are cross-coupled and applied to the gates of the p-type transistors M 3  and M 6  which control the supply of a higher voltage VDD 2  to the inverters  6  and  8 . If the input signal INPUT is at VDD 1  (ie. logic high) and the complementary input INPUT* is at ground, then transistors M 1 , M 5  and M 6  are on whilst transistors M 2 , M 3  and M 4  are off. Thus the output  13  of the shift circuit as a whole is at voltage VDD 2  (ie. logic high). The main limitation of this approach for large numbers of inputs is the need to supply two complementary input signals. 
     SUMMARY OF THE INVENTION 
     According to the invention there is provided a voltage level converter for converting an input signal at a first voltage level to an output signal at a second voltage level, the converter comprising an input for receiving said input signal, an output for outputting said output signal, a circuit node, precharge means for charging or discharging said circuit node to a third voltage level during a first time period by connection of said circuit node to a first voltage supply, isolation means for isolating said circuit node from said first voltage supply during a second time period, input means for changing the voltage at said circuit node in dependence on the voltage at said input during a third time period, and output means arranged so that the voltage at said output depends on the voltage at said circuit node. 
     Such a level converter does not require a complementary input signal, which is particularly advantageous if a number of such level converters are used in an arrangement having a large number of inputs. In addition, the isolation means can be used to prevent a direct current path from the first voltage supply to ground, thus reducing power consumption. 
     Said third voltage level may be substantially equal to said second voltage level. 
     This allows said first voltage supply to be used for both said output signal and for charging or discharging said circuit node. 
     Said third time period may coincide with, or lie within, said second time period. 
     The voltage level converter may further comprise an input node which is charged or discharged to a fourth voltage level during said first time period by connection of said input node to a second voltage supply. 
     Said isolation means may isolate said input node from said second voltage supply during said second time period. 
     Said second voltage supply may be negatively biased with respect to said input signal. 
     Said isolation means may comprise a first isolation field effect transistor the source and drain of which are connected between said second voltage supply and said input node. 
     Said isolation means may comprise a second isolation field effect transistor the source and drain of which are connected between said first voltage supply and said circuit node. 
     Said input means may comprise a sampler for sampling said input signal during said third time period. 
     Said sampler may comprise a sampling field effect transistor the gate of which is connected to a sampling signal. 
     The source and drain of said sampling field effect transistor may be connected between said input and said input node. 
     The voltage level converter may further comprise a latch for latching said output signal at the voltage determined by said input means. 
     The voltage level converter may further comprise a feedback field effect transistor, the source and drain of which are connected between said first voltage supply and said input node, and the gate of which is connected to said circuit node. 
     This provides the advantage of increased switching speed of the level converter, by increasing the switching speed of the input transistor mentioned below. 
     Said output means may comprise a CMOS inverter, the input of which is connected to said circuit node. 
     Said input means may comprise an input field effect transistor arranged to discharge said circuit node if said input signal is at a logic high level during said third time period. 
     The source and drain of said input field effect tristor may be connected between said circuit node and said second voltage supply, and the gate of said input field effect transistor may be connected to said input node. 
     The source of said input field effect transistor may be connected to said second voltage supply. 
     This provides the advantage that, if said second voltage supply is negatively biased with respect to said input signal, the drive efficacy of the input signal is increased. 
     The gate of said input field effect transistor may be connected to said negative voltage supply during said first time period, and isolated from said negative supply during said second time period. 
     Said input and output signals may be digital signals. 
     The invention also provides an active matrix liquid crystal display comprising a voltage level converter as described above. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Embodiments of the invention will now be described, by way of example only, with reference to the accompanying drawings, in which: 
     FIGS. 1 to  3  show three prior art level converters, as discussed above; 
     FIG. 4 shows a block diagram illustrating the basic components of an embodiment of the invention; 
     FIG. 5 shows a specific circuit level schematic of one embodiment of the invention; 
     FIG. 6 shows a simulation of voltages in the circuit of FIG. 5; and 
     FIGS. 7A and 7B are a schematic view of an active matrix display and associated data and scan drivers. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The basic components of an embodiment of the invention are shown in FIG. 4. A digital input signal  10 , varying between ground and a voltage VDD 1  (for logic levels 0 and 1 respectively) is converted to a digital output signal  12 , varying between ground and a higher voltage VDD 2 . Precharge and evaluate blocks  14  and  16  respectively are controlled by a control signal  18 . The precharge block  14  charges node A to voltage VDD 2  during a standby phase, by connecting node A to a voltage supply  20 . The precharge block  14  subsequently isolates node A from the voltage supply  20  during an active phase. 
     During the active phase, the input signal  10  is sampled by the evaluate block  16 . If the input signal  10  is high, then the evaluate block  16  discharges node A, and an inverter  22  produces a high output signal  12 , which is at the voltage VDD 2 . The evaluate block  16  and the inverter  22  are connected to a bias voltage VSS 1  which is negative with respect to the ground voltage (corresponding with the input signal logic level 0) mentioned above. The negative bias voltage VSSI increases the drive efficacy and switching speed of the evaluate block  16 . A latch  24  is also provided in order to latch the output signal  12  at a high level once node A has been discharged. 
     If the sampled input signal  10  is low, then the evaluate block  16  does not discharge node A, and the output signal  12  remains low (ie. at ground). 
     FIG. 5 illustrates one way of implementing the arrangement of FIG. 4 using seven field effect transistors M 1  to M 7  to form a digital data interface  30 . Transistor M 1  is an input sampling transistor controlled by a signal SAMPLE in order to sample an input signal INPUT. The input signal INPUT is a digital signal varying between ground and a low voltage level (eg. 3.3 volts) representing logic states 0 and 1 respectively. The digital data interface  30  of FIG. 5 converts the digital input signal to a digital output signal OUTPUT which varies between ground and a higher voltage level VDD, typically around 12 volts. Transistors M 5  and M 6  form a conventional CMOS inverter, of which node A forms the input Transistors M 3  and M 4  form a precharge—evaluate type inverter, and M 7  is a feedback transistor. M 7  can be regarded as forming part of the latch  24  of FIG.  4 . 
     The gates of NMOS transistor M 2  and PMOS transistor M 3  are respectively connected to a precharge signal PRECHARGE and a complementary signal PRECHARGE*. M 2  can be regarded as forming part of the precharge block  14  of FIG.  4 . 
     The sources of NMOS transistors M 2  and M 4  are connected to a negative source bias supply VB, of magnitude less than the threshold voltage of the n-type transistors M 2  and M 4 . 
     The digital data interface  30  of FIG. 5 operates as follows: 
     1. In a standby phase the PRECHARGE signal is high NMOS transistor M 2  discharges the gate of NMOS transistor M 4  (node B), while PMOS transistor M 3  charges the input (node A) of the output CMOS inverter (M 5 , M 6 ) up to voltage VDD. In this condition there is no conducting path from the supply voltage VDD to ground so quiescent power consumption is determined only by leakage currents in M 4  and M 5 . 
     2. In a subsequent active phase the PRECHARGE signal goes low, isolating node A from the power supply voltage VDD by switching off M 3 . Similarly node B is isolated from the negative supply voltage VB because M 2  is switched off. Nodes A and B therefore remain high and low respectively. 
     3. Next the SAMPLE signal goes momentarily high, switching on M 1  and charging node B to the voltage currently on the data input line  32 . The SAMPLE signal then goes low, isolating node B from the input. 
     4. If the sampled input signal is low, ie. OV, then the gate-source voltage across M 4  remains below threshold and so the input transistor M 4  does not conduct. Node A therefore remains at the precharged high voltage and the output remains low. 
     5. If the sampled input signal is high (e.g. 3.3V) then the gate-source voltage of the input transistor M 4  is equal to the logic high voltage (eg. 3.3V) minus the negative source bias voltage VB. Because this potential exceeds the threshold voltage of M 4 , M 4  begins to conduct. As M 4  discharges node A, the feedback transistor M 7  begins to turn on as a result of the dropping voltage at its gate, thus charging node B higher. This turns on M 4  harder and node A discharges more quickly. The final state of the circuit is with node B equal to VDD and node A equal to the source bias voltage VB. In this state there is no quiescent power consumption, apart from leakage effects, and the output is at VDD, ie. high. 
     The low voltage input signal is able to drive the output significantly faster than the prior art because the load transistor M 3  can only conduct in the standby state, and the input transistor M 4  therefore only has to conditionally discharge the capacitance at the output node A. The negative source bias voltage VB also increases the switching speed by creating a larger gate overdrive for the input transistor M 4 . 
     The use of a low input data voltage inherently reduces the power consumption resulting from the input signals driving parasitic capacitances. The power consumption of the interface  30  is also low because the dynamic technique avoids simultaneous conduction of the input and load transistors M 4  and M 3 . Furthermore, the DC power consumption is negligible in both the precharged state and the latched state. 
     The circuit requires only a small number of transistors and so is very area efficient. 
     The source bias input VB can be set to the optimum value for the threshold voltage of a given process. This flexibility is particularly important for TFT implementations which exhibit wide process variations. 
     FIG. 6 is a simulation of voltages in the circuit of FIG. 5, showing the timing of the relevant signals. From top to bottom, FIG. 6 shows the variation of the following voltages, PRECHARGE, PRECHARGE*, SAMPLE, INPUT, Node B, Node A, OUTPUT. The simulation uses an approximation for polysilicon thin film transistors (TFTs) with high threshold voltages of ±4V. The digital input level is 3.3V. It is clear from this simulation that the delay between the OUPUT high and the sample pulse is of the order of 15 nS, illustrating high speed operation, despite poor performance of the polysilicon TFTs. 
     FIGS. 7A and 7B illustrate an application of the invention to the field of monolithic digital data drivers for active matrix displays. An active matrix display  40  comprising a plurality of liquid crystal picture elements (LC pixels), one of which is shown enlarged and represented schematically at  42 , is controlled by a scan driver  44  and a plurality of data drivers  45  formed from a distributed controller  46 , a digital sampling array  48 , and plurality of digital-to-analogue converters  50 . In this type of display, both the drivers  45  and the active matrix  40  are fabricated using thin film transistors (TFTs), typically polysilicon. It is the combined function of the data line and scan line drivers  45  and  44  to provide analogue data voltages to the electrodes  52  of the LC pixels corresponding to the image data to be displayed. 
     In a typical digital data driver, the input video data is communicated in n-bit parallel RGB format, with n typically equal to 6 or 8 bits per colour. The display driver may therefore receive up to 24 (ie. 3×8) parallel digital data lines, and so low voltage operation is necessary to minimise power consumption from charging parasitic capacitance. This must be achieved despite the high threshold voltage of the TFTs in the driver. 
     In this application, each RGB column of the data driver  45  comprises n digital sampling circuits of the type shown in FIG. 5, to sample and store the parallel input data ready for subsequent digital-to-analogue conversion. The PRECHARGE and SAMPLE signals for the sampling circuits in each column are provided by means of the distributed controller  46 , which may be of the type disclosed in British Patent Application 9706941.3. In operation, the precharge signal is applied to a particular column while a previous column is sampling the data on the input lines. Once the sampling circuits are precharged, the precharge signal for that column goes low and the column is ready to sample the data lines. At the appropriate moment, when the data on the input lines corresponds to the particular column in question, the distributed controller  46  generates the sample pulse for that column and the data currently on the input lines is sampled and stored. The process is repeated for each column in the driver  45  until a line of data has been sampled. Once the data is stored, the conversion from digital to analogue format and subsequent data line charging can proceed in any suitable manner.