Abstract:
In integrated circuitry having both analog and digital circuits fabricated on the same substrate, switching transients produced by the digital circuitry can propagate through the substrate and induce deleterious effects in the associated analog circuitry. Such switching transients are greatly minimized by the disclosed family of CMOS logic circuits in which a constant DC bias current is steered to change logic states.

Description:
FIELD OF THE INVENTION 
     The present invention relates to integrated circuitry, and more particularly relates to a family of current-steering digital logic circuits that greatly minimizes switching current transients, thereby facilitating operation of digital and analog circuitry on common substrates. 
     BACKGROUND AND SUMMARY OF THE INVENTION 
     CMOS mixed-mode integrated circuits are a topic of increasing interest and importance. The present trend is towards mixed-mode monolithic systems containing higher frequency and higher accuracy analog circuitry combined with increasingly complex digital circuitry. Examples of such circuitry include 16-20 bit self-correcting successive-approximation converters and sigma-delta analog-to-digital (A/D) converters. The achievable accuracy of many such systems is now limited by the adverse effects of digital switching noise associated with conventional CMOS static logic circuitry. In the future, this problem will become proportionately worse at higher frequencies due to the unavoidable reduction in the power supply noise rejection capabilities of the analog circuitry at higher frequencies. 
     CMOS static logic is widely used because of its high packing densities, noise margins, and operating frequencies. Although a conventional CMOS logic gate ideally dissipates zero static power, a large overlap current pulse (˜1 mA/gate) flows from V dd  to GND during its state transition. Large current pulses flowing through the inductances and resistances associated with the substrate, power supply lines, bonding wires, package pins, etc., typically cause 1 volt or more of switching noise on the V dd  and GND lines. The accuracy of mixed-mode ASIC&#39;s is often limited by the coupling of the switching noise into the analog circuitry that shares the same substrate (FIG. 9). 
     One way to increase the accuracy of mixed-mode ASIC&#39;s is to increase the power supply noise rejection capabilities of the analog circuitry. With fully-differential analog architectures, power supply noise appears as a common-mode signal that is rejected by the common-mode-rejection-ratio (CMRR). However, CMRR usually decreases in direct proportion to increases in frequency. 
     A reduction in the transmission of digital switching noise between the digital and analog subsections may be obtained by using guardbanding and/or power supply separation techniques. Separate analog and digital power supply lines are used to minimize the common impedance. 
     Further success in reducing the transmission of switching noise between the analog and digital subsections is determined by the nature of the power supply connections to the substrate. When laying out digital cells using a p-well CMOS technology, it is common practice to incorporate within each cell several n +   ohmic contacts from the digital V dd  supply line to the substrate to deter latch-up. Hence, switching noise associated with digital V dd  is unavoidably coupled directly into the analog circuitry via the common substrate. 
     Significant increases in the accuracy of mixed-mode systems may also be achieved by reducing the amount of switching noise generated by the digital logic. The switching noise associated with conventional logic originated with the overlap current pulses generated during transitions. More precisely, the switching noise arises because the power supply currents are not constant. One way to implement logic circuits with constant supply currents is to use MOS differential pairs to switch constant current sources. Such a technique is disclosed in copending application Ser. No. 07/693,532 entitled MOS Folded Source-Coupled Logic. Although this approach achieves low power-supply noise, it requires several additional transistors per gate, and the MOSFETs are required to have large W/L ratios at high frequencies. 
     In accordance with the present invention, switching current transients in mixed mode circuitry are greatly minimized by use of source-coupled differential CMOS logic circuits wherein a constant DC bias current is steered to change logic states. Internal voltage swings are typically less than one volt. Consequently, measured power supply (V dd ) current spikes are typically only 15 μA for current steering inverter implemented in a 2 μm p-well CMOS technology--a reduction of two orders of magnitude compared to the 1.5 mA current spikes typical of a conventional static CMOS inverter. The reduction in digital switching noise allows the development of higher performance on-chip analog circuitry in CMOS mixed-mode applications. 
     The foregoing and additional features and advantages of the invention will be more readily apparent from the following detailed description, which proceeds with reference to the accompanying drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1A is a schematic diagram of a MOS inverter according to one embodiment of the present invention. 
     FIG. 1B is a schematic diagram of the MOS inverter of FIG. 1A adapted for a fanout of 2. 
     FIG. 2 is a graph depicting the DC transfer characteristic of the inverter of FIG. 1A as a function of applied input current. 
     FIG. 3 is a schematic diagram of a MOS two-input NOR gate according to another embodiment of the present invention. 
     FIG. 4 is a more detailed schematic diagram of the MOS inverter of FIG. 1A detailing one form of a suitable constant current source. 
     FIG. 5 is a graph depicting the power-delay performance of a logic circuit according to the present invention as a function of bias current with a V dd  voltage of 3 volts. 
     FIG. 6 is a graph like that of FIG. 5 but with a V dd  voltage of 5 volts. 
     FIG. 7 is a schematic diagram of an inverter like that of FIG. 1A but arranged in a voltage-mode topology. 
     FIG. 8 is a schematic diagram of a two-input NOR gate like that of FIG. 3 but arranged in a voltage-mode topology. 
     FIG. 9 shows an integrated circuit having both digital and analog circuitry on a common substrate. 
    
    
     DETAILED DESCRIPTION 
     Referring first to FIG. 1A, the operation of a first form of a current steering MOS (ISMOS) logic circuit 10 according to the present invention is premised on the use of currents, rather than voltages, to represent the logic signals. Assuming that the circuit 10 shown drives an identical stage, and that (W/L) 2  =2(W/L) 1 , then I in  =I gives I out  =0 and I in  =0 results in I out  =I. Hence, this circuit functions as a current-mode inverter without requiring a differential pair for current switching. Moreover, the total current drawn from the power supply is constant, I. The internal voltage swings associated with the current logic levels are typically designed to be 500 mV to 1 Volt. 
     One drawback of this implementation of ISMOS logic is that each fanout requires an identical output branch, as illustrated by the inverter 12 with fanout=2 shown in FIG. 1B. A DC transfer characteristic for the ISMOS inverter of FIG. 1A is shown in FIG. 2. It can easily be shown that equal high (NM H ) and low (NM L ) noise margins equal to I/2 are obtained by choosing (W/L) 2  =2(W/L) 1 . These ratios, of course, can be varied with corresponding changes in the noise margins. 
     A two-input (with fanout=1 capability) current-mode ISMOS NOR gate 14 is shown in FIG. 3, and a complete ISMOS inverter 10 is shown in FIG. 4. The constant current source load is here realized with a saturated PMOS device. V bias  is derived as part of a simple p-channel current-mirror circuit with bias current I. This form of ISMOS may be recognized to be somewhat reminiscent of bipolar integrated injection logic (I 2  L) as disclosed in Berger, H. et al, &#34;Merged-Transistor Logic (MTL)--A Low-Cost Bipolar Logic Concept,&#34; IEEE J. Solid-State Circuits, vol. SC-7, pp. 340-346, Oct. 1972, and Hart, K. et al, &#34;Integrated Injection Logic: A New Approach to LSI,&#34; IEEE J. Solid-State Circuits, vol. SC-7, pp. 346-351, Oct. 1972. 
     The power supply current spikes in ISMOS arise from two different mechanisms. Referring to FIG. 4, if driving an identical stage, as I out  varies from 0 to I, the output voltage varies from V T  to {V T  +[2I/k n  &#39;(W/L) 1   1/2  ]}. The total voltage swing, ΔV, is typically 0.5 to 1 Volt. As the output voltage varies, the bias current supplied by M 3  varies by an amount approximately equal to λ(ΔV)I due to the channel-length modulation effect on M 3 . The size of this supply current variation is typically about 0.03I for a 2 μm CMOS technology. The use of a cascode PMOS current-source load to eliminate this source of noise is under investigation. 
     The fundamental source of power supply current spikes in ISMOS is the displacement current (i=C dV out  /dt) associated with charging and discharging the `supply capacitances` associated with the output node. Supply capacitances are defined as the total capacitances between the output node and V dd  (C dd ) or GND (C GND ). The rate of change of the output voltage is approximately 
     
         dV.sub.out /dt=I/C.sub.out                                 (1) 
    
     where from FIG. 4, (driving an identical stage) 
     
         C.sub.out =C.sub.gs1 +C.sub.gb1 C.sub.db1 +C.sub.gs2 +C.sub.gb2 +C.sub.gd2 +C.sub.db2 +C.sub.db3 +C.sub.sb3.                         (2) 
    
     The displacement current spike into V dd  is given by 
     
         i.sub.dd =C.sub.dd dV.sub.out /dt=(C.sub.db3 /C.sub.out)I  (3) 
    
     and the current spike into GND is 
     
         i.sub.GND =C.sub.GND dV.sub.out /dt=(C.sub.gs1 +C.sub.gb1 +C.sub.db1 +C.sub.gs2 +C.sub.gb2 +C.sub.db2 /C.sub.out)I.            (4) 
    
     In a representative embodiment, a p-well CMOS technology is used, and therefore the substrate must be directly connected to the digital V dd  power supply to deter latch-up. Equations (3) and (4) suggest that V dd  current spikes can be minimized by connecting the constant current sources to V dd  as shown in FIG. 4. Although equation (4) predicts that the GND spikes may be relatively large, they flow only in the digital GND line which is not directly connected to the substrate in a p-well design. Therefore, GND noise does not adversely affect the analog circuitry by directly coupling through the substrate. 
     Since a constant current is switched in an ISMOS gate, it can be shown that for a given DC bias current, the speed of the gate is increased by decreasing the capacitance of the switching transistors. Hence, the performance of ISMOS will continue to improve as CMOS technology is scaled to smaller feature sizes. 
     A disadvantage of ISMOS in comparison to conventional CMOS logic is that it dissipates static power. However, the dynamic power (CΔV 2  f) of ISMOS is much smaller than that of conventional CMOS logic due to the much smaller logic swing. Hence, ISMOS actually dissipates less total power at higher frequencies than static logic. The bias current of ISMOS can be varied to minimize the total power dissipation for a given frequency of operation. 
     The efficacy of the disclosed technique has been verified by integrating several prototype circuits using MOSIS 2 μm p-well CMOS technology. An exemplary circuit is a 49-stage ring oscillator with various fanouts. 
     A key advantage of ISMOS is that it generates small power supply current spikes. The measured V dd  power supply current spikes in the fabricated ring oscillator show the square-wave component due to the channel-length modulation effects on the PMOS current source to be about 2 μA peak, and the glitch component due to the displacement currents flowing in the supply capacitances to be about 3 μA peak. The measured power-delay performance for the 49-stage ISMOS ring oscillator is shown in FIGS. 5 and 6 for V dd  =3.0 and 5.0 volts. At higher currents, the minimum propagation delay has been measured at about 500 pS for the ring oscillator with fanout=1. 
     From the foregoing, it will be recognized that ISMOS is well suited to high-precision high-speed mixed-mode ASIC applications. By steering a constant current from the V dd  power supply, the internal voltage swings are typically less than a volt. Hence, the power supply current spikes are only about 15 μA compared to more than a mA in a conventional static CMOS gate. The reduction in power supply current spikes translates to increased accuracy in the analog subsections of mixed-mode ASIC&#39;s. A minimum propagation delay of 500 pS with a power-delay product of 0.35 pJ was obtained early prototypes implemented in a 2 μm p-well CMOS technology. These prototypes have also been successfully tested using a 3 volt power supply. 
     Having described and illustrated the principles of our invention with reference to a preferred embodiment, it will be apparent that the invention can be modified in arrangement and detail without departing from such principles. For example, while the invention has been illustrated with reference to CMOS transistors, it will be recognized that the invention may find applicability with other fabrication technologies, some of which are not yet presently known. Similarly, while the invention has been illustrated with reference to representative logic circuits, namely an inverter and a two-input NOR gate, it will be recognized that the invention may similarly be applied to a variety of other arithmetic and logic circuits. Still further, while the invention has been particularly illustrated with reference to embodiments employing p-well CMOS technology, it should be apparent that transposition of device types can readily be achieved with only minor changes. 
     To illustrate some of the other forms the present invention may take, consider FIGS. 7 and 8. FIG. 7 shows an inverter akin to that of FIG. 1A. FIG. 8 shows a two-input NOR gate akin to that of FIG. 3. However, instead of using currents to represent different logic states (as were used in FIGS. 1A and 3), the topologies of FIGS. 7 and 8 permit voltages to represent different logic states. Such circuits may be termed voltage-mode ISMOS, as opposed to the current-mode circuits of FIGS. 1-4. 
     In view of the wide variety of embodiments to which the principles of our invention may be applied, it should be apparent that the detailed embodiments are illustrative only and should not be taken as limiting the scope of our invention. Rather, we claim as our invention all such modifications as may come within the scope and spirit of the following claims and equivalents thereto.