Abstract:
A dual supply bidirectional level shifter performs voltage level shifting in two directions, low to high and high to low. A feedback control branch and a control stage inverter are provided that reduce leakage power and allow for low delay time while also allowing for a small circuit footprint.

Description:
BACKGROUND OF THE INVENTION 
     The present invention relates to digital electronic circuits and, more particularly, to a voltage level shifter circuit that receives an input signal and generates an output signal that is a level-shifted version of the input signal. 
     Generally speaking, components and nodes in digital logic circuits transition from one logic level to another during the operation of the circuit. These transitions typically are between a logical high state at some voltage above ground level, and a logical low state at ground level. Occasionally, different circuits operating at different logical high voltage levels are required to interface with one another thereby requiring the voltage level of one circuit to be shifted with respect to the voltage level of the other circuit. For example, the voltage in one circuit may have a logic high voltage level of 0.75V and the voltage in the other circuit may have a logic high voltage level of 1.32V. The first circuit has a voltage swing of zero to 0.75V and the second circuit has a voltage swing of zero to 1.32V. Level shifters provide the connection between two such circuits, shifting the level of the signals from the first voltage swing to the second voltage swing. 
     One conventional voltage level shifter is illustrated in  FIG. 1 . This circuit attempts to eliminate static current, i.e. leakage current, consumption using a feedback circuit and a pull-device. The circuit comprises a first CMOS inverter  100  having complementary MOSFETs P 1  and N 1 , a second CMOS inverter  110  having complementary MOSFETs P 2  and N 2 , a third CMOS inverter  120  with complementary MOSFETs P 3  and N 3 , and a feedback unit  130  comprising P-type MOSFET P fb , N-type MOSFET N fb  and pull up device P pu    140 . 
     Each of the first and second inverters  100 ,  110  receives an input signal at a voltage V in  at a first, lower voltage level V DDL . The first inverter  100  outputs the inverse of the input signal at a node  150 , which is input to the third inverter  120 . The third inverter  120  inverts the signal at node  150  and provides an output signal V out  at voltage level V DDH , where the logical state of V out  reflects that of V in . Thus, V out  is a level-shifted version of V in . 
     The output of the second inverter  110  also is the inverse of the input voltage V in , at the lower voltage level V DDL . This is fed to the gate of device N fb  of feedback unit  130 . Similarly, the voltage at node  150  is fed to the gate of the device P fb  of the feedback unit  130 . The feedback unit  130  provides an output signal at node  170  that is used to drive the gate of pull-up device P pu    140 . 
     Typically, the digital voltage level shifter of  FIG. 1  operates as the input voltage at V in  transitions between a logical high at the first voltage V DDL  and a logical low at ground voltage where it is desired that the output voltage V out  reflects the logical state of V in , but at the level shifted voltage, V DDH . In this circuit, the input voltage V in  is at a first, lower voltage V DDL  and the output voltage is at a second, higher voltage V DDH . It is believed that improvements may be realized in reduction of leakage current when V IN  rises to a logic high of V DDL . For instance, and analyzing the case when V IN  rises to a logic high of V DRL , N 1  turns on, but P 1  initially does not completely turn off since the source of P 1  is at a voltage level of V DDH . Thus, static current temporarily flows through P PU , P 1 , and N 1 . Given the nature of normal CMOS processes, N-channel FETs have approximately twice the current sinking and sourcing capability of identically-sized P-channel FETs. Additionally, the circuit of  FIG. 1  has two P-channel FETs, P 1  and P PU , connected in series, thereby further reducing the strength of P 1  and P PO  in comparison to N 1 . Therefore, N 1  succeeds in pulling node  150  to ground. V IN  also turns N 2  on and P 2  completely off (since the source of P 2  is attached to V DDL ), thus pulling node  160  to ground. With the gates of both P FB  and N FB  pulled low, node  170  is pulled up to V DDH  volts, thereby shutting off P PU  and eliminating the static current that previously flowed through P PU , P 1 , and N 1 , and terminating the drive fight between P 1  and N 1 . Also, with node  150  being at ground, P 3  is on, N 3  is off, and V OUT  is pulled up to V DDH , all in response to V IN  rising to V DDL . 
     As V IN  transitions to logical low, P 1  goes on and N 1  goes off, however, node  150  remains at logical low (ground) because in the previous cycle of operation (as described above), P pu  was switched off, thereby isolating node  150  from V DDH , at least temporarily. 
     Additionally, P 2  goes on and N 2  goes off, thereby pulling node  160  to logical high at V DDL . In turn, N fb  goes on and since node  150  is currently at ground, P fb  is on, which means a leakage path exists from V DDH  to ground through P fb  and N fb . Node  170  is being driven by P fb  to be pulled up to V DDH  and by N fb  to be pulled to ground. Because N fb  is of a physically larger size in order to influence the voltage at node  170  as described above, the larger-sized device N fb  wins the drive fight eventually pulling node  170  to ground. Only then is node  150  pulled up to V DDH  through P pu  and P 1 , thereby switching P fb  off and cutting off the leakage from V DDH  through P fb  and N fb  to ground. 
     The above-described circuit operation is less than optimal because of the leakage current from V DDH  to ground through P fb  and N fb , which flows for a relatively long time which, in turn, requires the physical size of device N fb  to be relatively large. Additionally, the circuit of  FIG. 1  takes a relatively long time for the output voltage to transition to ground in response to a corresponding transition on the input voltage. Accordingly, it would be advantageous to have a digital voltage level shifter that provides some improvement on the above problems. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The present invention is illustrated by way of example and is not limited by the accompanying figures, in which like references indicate similar elements. Elements in the figures are illustrated for simplicity and clarity and have not necessarily been drawn to scale. 
         FIG. 1  illustrates a conventional digital voltage level shifter; and 
         FIG. 2  illustrates a digital voltage level shifter in accordance with an embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     In an embodiment of the invention there is provided a digital voltage level shifter for receiving an input signal of a first voltage swing and outputting an output signal of a second voltage swing, the output signal being a level-shifted version of the input signal, the digital voltage level shifter comprising: a first inverter comprising: a first input for receiving the input signal; and a first output; a second inverter comprising: a second input connected to the first output; and a second output for outputting the output signal; and a control stage for controlling switching of the digital voltage level shifter, the control stage comprising a feedback circuit branch having a control stage switch configured to assume a non-conducting state dependent on a logical state of the output signal. 
     In another embodiment of the invention there is provided a digital voltage level shifter for receiving an input signal of a first voltage swing and outputting an output signal of a second voltage swing, the output signal being a level-shifted version of the input signal, the digital voltage level shifter comprising: a first inverter comprising: a first input for receiving the input signal; and a first output; a second inverter comprising: a second input connected to the first output; and a second output for outputting the output signal; and a control stage for controlling switching of the digital voltage level shifter, the control stage comprising: a control stage inverter having a control stage inverter input for receiving the input signal; and a control stage inverter output; and wherein the second inverter comprises a second inverter switch having a switch control input controlled by the control stage inverter output. 
     In a further embodiment of the invention there is provided a method of operating a digital voltage level shifter configured to receive an input signal of a first voltage swing and output an output signal of a second voltage swing, the output signal being a level-shifted version of the input signal, the method comprising: providing a first inverter comprising: a first input for receiving the input signal; and a first output; providing a second inverter comprising: a second input connected to the first output; and a second output for outputting the output signal; providing a control stage for controlling switching off the digital voltage level shifter, the control stage comprising a feedback circuit branch having a control stage switch; applying an input signal to the first input; and controlling the control stage switch to assume a non-conducting state dependent on a logical state of the output signal. 
     Embodiments of the invention may provide significant technical benefits in comparison with conventional techniques. In the first place, the leakage current that flows for a relatively long time in the conventional circuit of  FIG. 1  from V DDH  to ground through P fb  and N fb , is substantially reduced, as the digital voltage level shifter may be configured for the control stage switch to assume a non-conducting state dependent on a logical state of the output signal. Secondly, the physical size of the switching device that corresponds to device N fb  in the conventional circuit of  FIG. 1  may be of a significantly reduced size, thereby leading to a reduced footprint for the circuit. Thirdly, significant improvements may be realized in the time taken for the output voltage to fall to logical low/ground in response to the input voltage going to logical low. 
     The terms “a” or “an,” as used herein, are defined as one or more than one. Also, the use of introductory phrases such as “at least one” and “one or more” in the claims should not be construed to imply that the introduction of another claim element by the indefinite articles “a” or “an” limits any particular claim containing such introduced claim element to inventions containing only one such element, even when the same claim includes the introductory phrases “one or more” or “at least one” and indefinite articles such as “a” or “an.” The same holds true for the use of definite articles. 
     Unless stated otherwise, terms such as “first” and “second” are used to distinguish between the elements such terms describe. Thus, these terms are not necessarily intended to indicate temporal or other prioritization of such elements. 
     Because the apparatus implementing the present invention is, for the most part, composed of electronic components known to those skilled in the art, full details will not be explained in any greater extent than that considered necessary for the understanding and appreciation of the underlying concepts of the present invention and in order not to obfuscate or distract from the teachings of the present invention. 
     Referring now to  FIG. 2 , a digital voltage level shifter  200  in accordance with an embodiment of the invention is illustrated. The digital voltage level shifter  200  includes a voltage translation stage  202 , a driver stage  204  and a control stage  206 . The voltage translation stage  202  receives an input voltage and translates the input voltage to the voltage to which it is to be level shifted. The driver stage  204  drives the level-shifted output voltage. The control stage  206  controls switching of the digital voltage level shifter  200 . 
     The voltage translation stage  202  comprises a first inverter  208  comprising a complementary CMOS pair of P-type MOSFET M 5   210  and N-type MOSFET M 6   212 . The first inverter  208  has a first input  214  for receiving the input signal which is at a first or lower voltage level V ddi . The first inverter  208  has a first output  216  that outputs a first output signal at node y, which is the logical inverse of the first input  214 , but translated to the second, higher voltage level V ddo . The drain terminals of M 5   210  and M 6   212  are tied together to provide the first output  216  of first inverter  208 . 
     The voltage translation stage  202  further comprises another switching device M 4   218  which, in the embodiment of  FIG. 2 , is a P-type MOSFET having a drive input  220  at its gate. The source of M 4   218  is driven by the second, higher-voltage supply V ddo , while the drain of M 4   218  is connected in series to the source of M 5   210 . 
     The driver stage  204  comprises a second inverter  222  comprising a P-type MOSFET M 7   224  and N-type MOSFET M 8   226 . Note that in the embodiment of  FIG. 2 , the second inverter  222  is not a true complementary CMOS inverter since the gates of M 7   224  and M 8   226  are not tied together. Instead, the gate of M 8   226  has a switch control input  228  driven by an output of the control stage  206 , which will be described in further detail below. On the other hand, the gate  230  of M 7   224  is driven by the output of the first inverter  208  on node y. The gate of M 7   224  is the second input of the second inverter  222 . The drains of M 7   224  and M 8   226  are tied together and provide the second output  232 , which is the output of the digital voltage level shifter  200 . Additionally, the source of M 7   224  is tied to the higher-voltage supply V ddo , and the source of M 8   226  is tied to ground. 
     The control stage  206  comprises a feedback circuit branch  234  including three switches PMOSFET M 1   236 , NMOSFET M 2  (control stage switch)  238 , and NMOSFET M 3   240 . The first switch M 1   236  has a source connected to the higher-voltage V ddo  and a drain tied to a drain of the second switch M 2   238 . The second or control stage switch M 2   238  has a source connected to a drain of the third switch M 3   240 , and a source of the third switch M 3   240  is tied to ground. Thus, the control stage switch M 2   238  is connected in series between PMOS and NMOS transistors  236 ,  240 . In the embodiment of  FIG. 2 , the control stage switch M 2   238  is a low-voltage threshold device. 
     The first switch M 1   236  has a switch control input  242 , in which a gate of the first switch M 1   236  is connected to the output of the first inverter  208  at node y. The control stage switch M 2   238  has a switch control input  244 , in which a gate of the control stage switch M 2   238  is connected to the output of the second inverter  222 . The third switch M 3   240  has a switch control input  246 , in which a gate of the third switch M 3   240  is connected to an output of a third inverter  248 , as will be described below. 
     The control stage  206  further comprises a control stage inverter  248  comprising a complementary CMOS pair of PMOSFET M 9   250  having a source connected to a lower-voltage supply V ddi , and NMOSFET M 10   252  having a source terminal tied to ground. The drains of M 9   250  and M 10   252  are connected together and provide the control stage inverter output  256  at node Ab, which drives device M 3   240  at its gate  246 , as noted above. Also the gates of M 9   250  and M 10   252  are tied together and form the control stage input  254  of the control stage inverter  248 . For the avoidance of leakage between V ddi  and ground through M 9   250  and M 10   252 , the gates M 9   250  and M 10   252  are driven by a signal A which has a range from ground to V ddi . Thus, the feedback circuit branch  234  has the third switch M 3   240 , which has its gate  246  controlled by the output  256  of the control stage inverter  248 . 
     In operation, when input A goes to logical high/V ddi , MOSFET device M 5   210  of the first inverter  208  is switched partially on, and transitions to being switched fully off after node net 131  is pulled up to logical high. This is driven by M 1   236  and M 2   238 , the gates y  242  and Z  244  of which are, respectively, at logical low and logical high/V ddo  and MOSFET device M 6   212  being switched on. This pulls the first output  216  of the first inverter  208  at node y to logical low/ground. In turn, this switches M 7   224  on and the second output  232  at node Z is pulled to high/V ddo . So, the logical state of the second output at node Z reflects the logical state of input A, but at the higher voltage level, V ddo . 
     Because the first input A on  214  is at logical high/V ddi , and because the control stage inverter input A  254  is tied to the first input  214 , PMOSFET M 9   250  of the control stage inverter  248  is switched off, corresponding NMOSFET M 10   252  is switched on and the control stage inverter output  256  at node Ab is pulled to logical low/ground. This drives second inverter switch M 8   226  to off, as the switch control input  228  of M 8   226  is tied to the control stage inverter output  256  at node Ab. This ensures that the second output  232  at node Z remains at logical high/V ddo . 
     As node y is at logical low/ground, switch M 1   236  is turned on because its gate is tied to node y  216 . Additionally, and because the second output Z  232  is tied to the switch control input  244  of control stage switch M 2   244 , this device switches on. Further, the third switch NMOSFET M 3   240  of the feedback circuit branch  234  is also switched off because it is driven by the control stage inverter  248  output Ab  256 . As a consequence, the node net 131   220  at the gate of M 4   218  is pulled up approximately to logical high/V ddo . Actually, the gate of M 4   218  is at the level of V ddo  less the threshold voltage V t1  of the control stage switch M 2   238  which is, in this embodiment, a low voltage threshold device. Therefore, this means that M 4   218  is eventually switched off, as the difference between the gate voltage of M 4   218  and the source voltage should be greater than the threshold voltage V t  of M 4   218 , in this embodiment a standard threshold voltage device. That is, (Vddo−Vt 1 ) gate voltage of m 4 −(Vdd 0 ) source voltage of m 4 |=|Vt 1 |&lt;|Vt| of M 4   218  as the standard MOSFET threshold voltage is greater than the threshold voltage of a LVT MOSFET. As a consequence of M 4   218  being switched off, there is no leakage current from V ddo  to ground through M 4   218 , M 5   210  and M 6   212 . 
     Now, considering the case where input A  214  goes to logical low/ground, M 6   212  is switched off and M 5   210  is switched on. However, because M 4   218  remains in the off state from the previous cycle of operation when the input voltage switched to a logical level high, as noted above, the first output  216  of the first inverter  208  at node y is floating. 
     Additionally, M 9   250  is switched on and M 10   252  is switched off, thereby pulling up the control stage inverter output  256  at node Ab to logical high/V ddi . In turn, this switches on M 8   226 , pulling the second output  232  at node Z to ground. (M 7   224  is in the conducting state at this time, but only for a very short period. M 3   240  and M 8   226  are switched on simultaneously, so net 131   220  is pulled to ground, in turn switching on M 4   218  and pulling node y  216  to logical high/V ddo  very quickly through M 4   218  and M 5   210  and, therefore, M 7   224  is switched off.) Therefore, the logical state of the second output  232  at node Z reflects the logical state of the signal on input  214 , on node A. The control stage inverter  248  has a control stage inverter input  254  for receiving the input signal A and a control stage inverter output  256 . The second inverter switch  226  of the second inverter  222  has a switch control input  228  controlled by the output Ab  256  of the control stage inverter  248 . 
     It will also be appreciated that  FIG. 2  illustrates a digital voltage level shifter  200  for receiving an input signal of a first voltage swing and outputting an output signal of a second voltage swing, the output signal being a level-shifted version of the input signal. In the embodiment of  FIG. 2 , the switch control input  228  of the second inverter  222  is connected to the control stage inverter output  256  on node Ab. 
     Immediately following the switching on of M 8   226 , which pulls the second output  232  at node Z to ground, the control stage switch M 2   238  assumes a non-conducting state because its gate  244  is tied to and driven by output  232  at node Z, and the feedback circuit branch  234  is put into an open-circuit condition, regardless of the state of the first switch M 1   236  (still on at this point from the previous cycle of operation when the input on  214  went to logical high) and the third switch M 3   240 . 
     Further, because the control stage inverter output  256  has been pulled to logical high/V ddi , then M 3   240  is also switched on, thereby pulling the voltage at node net 131   220  to ground, switching on M 4   218  and driving the voltage of the first output  216  of the first inverter  208  at node y to logical high/V ddo , thereby switching off M 7   224 . Switching on of M 8   226  as mentioned above pulls the second output  232  at node Z to ground. So, the second inverter switch M 8   226  (an NMOSFET) is configured to assume a conducting state when the input signal A is at a logical low state. The second inverter  222  further comprises a PMOSFET M 7   224 , which has a gate terminal that receives the second input  230  of the second inverter  222  at node y. 
     After the first output  216  at node y has been pulled up to logical high, the first switch M 1   236  switches off because its gate  242  is tied to, and driven by, node y. However, and as mentioned above, the feedback circuit branch  234  is already in open circuit due to the fact that the control stage switch M 2   238  is off because the output  232  at node Z has been driven low. 
     Thus, this provides a significant improvement from the conventional circuit of  FIG. 1 , because the gate  244  of the control stage switch M 2   238  is tied to node Z, as soon as the second output  232  at node Z goes to logical low, then, the control stage switch M 2   238  is immediately switched off, thereby cutting off any leakage current that might otherwise flow from V ddo  to ground through the first and third switches M 1   236  and M 3   240 , as was the case with the conventional circuit of  FIG. 1 . As discussed above, in the absence of the control stage switch M 2   238  and it being driven by the output Z, leakage current flow in the feedback circuit branch  234  is not cut out until the first output  216  at node y is driven high, thereby switching of the first switch M 1   236 . Consequently, the current rating and/or physical size of the third switch M 3   240  can be significantly reduced in comparison with device N fb  of  FIG. 1 . Therefore, the overall footprint of the digital voltage level shifter  200  may be significantly smaller than that of the conventional digital voltage level shifter of  FIG. 1 . 
     In the embodiment of  FIG. 2 , the concept implemented is that of using a single path from the higher-voltage supply Vddo to charge or discharge the intermediate nodes. This single path is cut off when the input assumes a low voltage domain high logic. The same circuit path and then re-made when the input voltage is at logical low, also reducing the fall delay, and assisted by an inverted input signal. 
     In the embodiment of  FIG. 2 , the control stage switch M 2   238  assumes a non-conducting state when the output signal of the second output  232  is in a logical low state. The control stage switch M 2   238  has a switch control input  244  connected to the second output  232  of the second inverter. As described above, in this embodiment, the control stage switch M 2   238  is an NMOSFET having its gate at the switch control input  244 , which is connected to the output of the second inverter at node Z so when the second output (node Z) goes low, the control stage switch M 2   238  goes to the non-conducting or off state. 
     The first switch M 1   236  also assists in helping to curtail leakage power in instances where the control stage switch M 2   238  is a LVT device; that is, in this embodiment, the control stage switch M 2   238  comprises a low voltage threshold (LVT) NMOSFET. Such devices have a higher sub-threshold leakage even when the voltage on the input (gate) of this transistor is sitting very close to, but not quite at, V ddo . Thus, when the first switch M 1   236  is in a non-conducting state, this helps prevent the possibility of breakdown of the control stage switch M 2   238  when it is a LVT device. Additionally, the combination of voltages at the inputs of the switches M 1   236 , M 2   238 , and M 3   240  that takes the voltage at node net 131   220  quickly to ground. 
     As noted above, the embodiment of  FIG. 2  may provide significant technical benefits when compared to, for example, the conventional circuit of  FIG. 1 . In this regard, circuit simulations were performed for the purposes of comparison between the conventional circuit  FIG. 1  and the level shifter  200  of  FIG. 2 . The simulations were conducted with the 55 nm technology mode, in the voltage range of 0.9V to 1.32V, with the models of best, worst, typ, bpwn and wnwp. Rise delay, fall delay and leakage were measured, yielding the results shown in Tables 1-4. 
     
       
         
               
             
               
               
               
               
               
               
             
               
               
               
               
               
               
             
           
               
                 TABLE 1 
               
             
             
               
                   
               
               
                 Transition times for low to high (0.9 V to 1.32 V) 
               
             
          
           
               
                   
                 Circuit 
                 Corner 
                 Temp C. 
                 Cell Rise (ns) 
                 Cell Fall (ns) 
               
               
                   
                   
               
             
          
           
               
                   
                 FIG. 2 
                 wcs 
                 −40 
                 0.111 
                 0.383 
               
               
                   
                 FIG. 1 
                 wcs 
                 −40 
                 0.119 
                 6.700 
               
               
                   
                 FIG. 2 
                 typ 
                 25 
                 0.093 
                 0.264 
               
               
                   
                 FIG. 1 
                 typ 
                 25 
                 0.100 
                 0.364 
               
               
                   
                   
               
             
          
         
       
     
     
       
         
               
             
               
               
               
               
               
             
           
               
                 TABLE 2 
               
             
             
               
                   
               
               
                 Leakage current for low to high (1.32 V to 0.9 V) 
               
             
          
           
               
                   
                 Circuit 
                 Corner 
                 Temp C. 
                 Leakage (W) 
               
               
                   
                   
               
               
                   
                 FIG. 2 
                 wcs 
                 150 
                 8.59E−08 
               
               
                   
                 FIG. 1 
                 wcs 
                 150 
                 1.36E−07 
               
               
                   
                   
               
             
          
         
       
     
     
       
         
               
             
               
               
               
               
               
               
             
               
               
               
               
               
               
             
           
               
                 TABLE 3 
               
             
             
               
                   
               
               
                 Input transition for high to low (1.32 V to 0.9 V) 
               
             
          
           
               
                   
                 Circuit 
                 Corner 
                 Temp C. 
                 Cell Rise (ns) 
                 Cell Fall (ns) 
               
               
                   
                   
               
             
          
           
               
                   
                 FIG. 2 
                 wcs 
                 −40 
                 0.113 
                 0.106 
               
               
                   
                 FIG. 1 
                 wcs 
                 −40 
                 0.115 
                 0.509 
               
               
                   
                 FIG. 2 
                 typ 
                 25 
                 0.093 
                 0.091 
               
               
                   
                 FIG. 1 
                 typ 
                 25 
                 0.095 
                 0.362 
               
               
                   
                   
               
             
          
         
       
     
     
       
         
               
             
               
               
               
               
               
             
           
               
                 TABLE 4 
               
             
             
               
                   
               
               
                 Leakage current for high to low (1.32 V to 0.9 V) 
               
             
          
           
               
                   
                 Circuit 
                 Corner 
                 Temp C. 
                 Leakage (W) 
               
               
                   
                   
               
               
                   
                 FIG. 2 
                 wcs 
                 150 
                 4.05E−08 
               
               
                   
                 FIG. 1 
                 wcs 
                 150 
                 5.92E−08 
               
               
                   
                   
               
             
          
         
       
     
     In tables 1-4, bcs refers to the best condition on which the chip is working i.e. chip is experiencing the most favorable conditions or more specifically when PMOS and NMOS are performing best means fastest (MOS behavior under these conditions). wcs refers to the worst condition on which the chip is working i.e. chip is experiencing the most unfavorable conditions or more specifically when PMOS and NMOS are performing worst means slowest. Typ refers to the typical condition on which the chip is working i.e. chip is experiencing the normal conditions or more specifically when PMOS and NMOS are performing typically means expected. bpwn refers to the best PMOS and worst NMOS condition on which the chip is working i.e. chip is experiencing the corner conditions or more specifically when PMOS are operating under the best conditions and NMOS are working under worst conditions. bnwp refers to the best NMOS and worst PMOS condition on which the chip is working i.e. chip is experiencing the corner conditions or more specifically when NMOS are working under best conditions and PMOS are working under worst conditions. 
     Table 1 illustrates the advantage of the level shifter  200  when compared with the level shifter of  FIG. 1  in the cell fall time, when the input signal is at 0.9V and being shifted to 1.32V at the output. Table 2 illustrates that the level shifter  200  has lower leakage power than the circuit of  FIG. 1 , when the input signal is at 0.9V and being shifted to 1.32V at the output. Table 3 illustrates that the level shifter  200  has faster cell fall time than the circuit of  FIG. 1  when the input signal is at 1.32V and shifting it to 0.9V at the output. Table 4 illustrates that the level shifter  200  has improved leakage power consumption over the circuit of  FIG. 1  when the input signal is at 1.32V and being shifted to 0.9V at the output. 
     Therefore, it can be seen that significant improvements are realized in the cell fall times. For instance, the simulation cell fall time when transitioning from logical low to logical high for the circuit of  FIG. 1  on the wcs process corner is 6.7 ns, compared with 0.383 ns as simulated for the circuit of  FIG. 2 . Further, the simulated leakage power loss is reduced from 1.36E-07 W in the circuit of  FIG. 1  to 8.59E-08 W of  FIG. 2 . In low to high voltage shifting, the circuit of  FIG. 2  can translate signals in the range of 0.75 V to 1.32 V under the bcs, typ, wcs, wnbp, and bnwp process corners. Further, signal translation by the level shifter  200  is bidirectional. The embodiment may be implemented in applications where shifting from a lower to a higher voltage is required, and also when shifting from a higher to a lower voltage. 
     Table 5 illustrates the FMEA (Failure Mode Analysis) results for the level shifter  200 . 
     
       
         
               
             
               
               
               
               
               
               
             
               
               
               
               
               
               
             
           
               
                 TABLE 5 
               
             
             
               
                   
               
               
                 FMEA RESULTS 
               
             
          
           
               
                   
                   
                   
                 Functional  
                 Cell Rise 
                 Cell Fall 
               
               
                 Corner  
                 Temp 
                 Voltage V 
                 result 
                 (ns) 
                 (ns) 
               
               
                   
               
             
          
           
               
                 wcs 
                 −40 
                 Vddi = 0.75, 
                 FAIL 
                 0.173 
                 1.06 
               
               
                   
                   
                 Vddo = 1.32 
                   
                   
                   
               
               
                 wcs 
                 25 
                 Vddi = 0.75, 
                 PASS 
                 0.173 
                 0.853 
               
               
                   
                   
                 Vddo = 1.32 
                   
                   
                   
               
               
                 wcs 
                 150 
                 Vddi = 0.75, 
                 PASS 
                 0.168 
                 0.670 
               
               
                   
                   
                 Vddo = 1.32 
                   
                   
                   
               
               
                 wcs 
                 −40 
                 Vddi = 0.76, 
                 PASS 
                 0.162 
                 0.986 
               
               
                   
                   
                 Vddo = 1.32 
                   
                   
                   
               
               
                 wcs 
                 25 
                 Vddi = 0.76, 
                 PASS 
                 0.166 
                 0.792 
               
               
                   
                   
                 Vddo = 1.32 
                   
                   
                   
               
               
                 wcs 
                 150 
                 Vddi = 0.76,  
                 PASS 
                 0.163 
                 0.637 
               
               
                   
                   
                 Vddo = 1.32 
                   
                   
                   
               
               
                   
               
             
          
         
       
     
     The results in the above tables demonstrate a significant reduction in the fall delay, approximately 20 times less when compared with the circuit of  FIG. 1 . Further, leakage is less when compared to the prior art, an improvement of between 50 and 100%, as may be derived from Tables 1 to 4. 
     Furthermore, this allows a significant area reduction in the footprint of the digital voltage level shifter in comparison to the prior art. Yet further, the circuit of  FIG. 2  is able to operate with an input voltage of down to 0.75 V. 
     By now it should be appreciated that there has been provided novel techniques for digital voltage level shifting which may be implemented across, for example, all low-power System on Chip (SoC) designs using multiple voltage domains, where it is required to pass signals from one voltage domain to another. This may be realized by having a switching device in a feedback circuit branch assume a non-conducting state when the output of the digital voltage level shifter falls to logical low. 
     Although the invention is described herein with reference to specific embodiments, various modifications and changes can be made without departing from the scope of the present invention as set forth in the claims below. Accordingly, the specification and figures are to be regarded in an illustrative rather than a restrictive sense, and all such modifications are intended to be included within the scope of the present invention. Any benefits, advantages, or solutions to problems that are described herein with regard to specific embodiments are not intended to be construed as a critical, required, or essential feature or element of any or all the claims.