Abstract:
A method for generating minimum shift keying (MSK) modulation signals with relatively reduced energy sidelobes relative to known methods. In one embodiment, a second derivative of the phase waveform is taken and convolved with a selectable smoothing function and integrated twice to generate a smooth phase waveform S2D --  MSK (i)=where .o slashed. (i)=π i d(i)/2T. In the alternate embodiment, the sidelobe energy can be varied by simply changing the smoothing function, which can be stored in a lookup table. In the embodiment, the MSK modulation signal has a constant amplitude envelope allowing it to be transmitted using class C amplifiers. Since the embodiment provides continuous phase derivatives, the frequency and thus the sidelobe energy is sufficiently reduced to minimize interference with cosite equipment and nearby communication channels bands.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a method for generating minimum shift keying (MSK) modulation signals and more particularly to a method for generating MSK modulation signals with reduced sidelobe energy relative to known MSK modulation techniques to reduce interference with colocated receiving equipment (cosite) and to reduce interference with adjacent channels in communication systems, such as single channel per carrier (SCPC) satellite links. 
     2. Description of the Prior Art 
     Methods for generating minimum shift keying (MSK) modulation signals are generally known in the art. Examples of apparatus for generating such MSK modulation signals are disclosed in U.S. Pat. Nos. 3,993,868; 4,516,087; 4,897,629 and 5,020,079. Known apparatus for decoding MSK modulation signals are disclosed in U.S. Pat. Nos. 4,942,592 and 5,170,131. 
     A baseband minimum shift keying (MSK) modulation signal is comprised of an in-phase component (I) or channel and a quadrature component or channel (Q). The in-phase component (I) and quadrature component (Q) are time offset from one another by 90°, such that the vector sum of the two components (I and Q) results a constant amplitude envelope, which can therefore be amplified, for example, by way of a non-linear class C amplifier. 
     Both the in-phase (I) and quadrature phase (Q) component signals are formed from half cycle sinusoidal waveforms of varying polarity, depending on the bit value being either a &#34;0&#34; or a &#34;1&#34;. Even numbered bits are modulated by the in-phase component signal while odd numbered bits are generally modulated by the quadrature component signal (Q). When successive alternate bits with the same phase value are modulated, the sinusoidal waveforms will have the same polarity resulting in a relatively sharp discontinuity in the phase waveform. Such a discontinuity in the phase waveform results in high frequency components due to the non-continuous phase derivative which, in turn, results in spectrum spreading. 
     Various techniques are known for minimizing the spectral spreading in such a situation, for example, as disclosed in U.S. Pat. Nos. 4,516,087 and 4,897,620, hereby incorporated by reference. In the &#39;620 patent, during a condition when successive alternate bits are to be modulated, a continuous transition modulation signal is utilized to maintain the continuity of the component signal waveform, (I or Q). As such, the first derivative of the phase waveform (i.e. frequency) during such sharp phase transitions will be continuous. In order to maintain a constant envelope amplitude, the other component signal (I or Q) is adjusted. 
     The technique disclosed in the &#39;087 patent utilizes a predetermined number of stored modulation waveform patterns, selected as a function of the values of the successive alternate data bits for either the in-phase channel (I) and the quadrature phase channel (Q). The stored waveforms are used to smooth transitions of the phase waveform whenever the successive alternate data bits for either of the channels have the same value in order to prevent discontinuities. The other component signal is adjusted so that the sum of the component signals is constant. 
     While the techniques disclosed in the &#39;087 and &#39;620 patents reduce the spectral spreading of the MSK modulation signal and in particular reduce the sidelobe energy, such techniques are not suitable for use in all applications. For example, transmission of MSK modulation signals can interfere with colocated receiving equipment (cosite) or adjacent channels in the case of a single channel per carrier (SCPC) satellite links. 
     Cosite interference, for example, in a joint tactical information distribution system (JTIDS) application, arises when the transmit and receive antennas are sufficiently coupled and measures are not taken to filter the offending out-of-band transmitted spectrum. Interference can also occur over satellite channels in a single channel per carrier (SCPC) where out-of-band emissions are severely restricted by law. 
     SUMMARY OF THE THE INVENTION 
     It is an object of the present invention to solve various problems in the prior art. 
     It is yet another object of the present invention to provide a method for generating minimum shift keying (MSK) modulation signals with reduced sidelobe energy for minimizing the interference with colocated receiving equipment. 
     It is yet another object of the present invention to provide a method for generating MSK modulation signals which minimize interference with adjacent communication channels. 
     Briefly, the present invention relates to a method for generating minimum shift keying (MSK) modulation signals with relatively reduced energy sidelobes relative to known methods. In one embodiment of the invention, discontinuities in the baseband phase modulation signal are eliminated by utilizing a continuous sinusoidal MSK modulation signal; MSK (t)=exp jπtd(i)/2T, where ##EQU1## t is time and d(i)=±1 for a period T in an interval I. The time varying frequency term, A sin (πt/T)/2 results in smooth phase waveform with a continuous first derivative (i.e. continuous frequencies). In an alternate embodiment, a second derivative of the phase waveform is taken and convolved with a selectable smoothing function and integrated twice to generate a smooth phase waveform S2D --  MSK (t)=e j .o slashed.(t), where ##EQU2## where φ&#34;(t)dt=d(i),μ 0  (t 1 )h(t) and μ 0  (t) is an impulse function and h(t) is a smoothing function. 
     --denotes convolution 
     In the alternate embodiment, the sidelobe energy can be varied by simply changing the smoothing function, which can be stored in a lookup table. In this embodiment, the baseband complex modulation representation of the signal has a constant amplitude envelope allowing it to be transmitted using class C amplifiers. Since the embodiment ensures continuous phase derivative, the sidelobe energy being significantly reduced minimizes interference with other cosite equipment and nearby communication channels bands. 
    
    
     DESCRIPTION OF THE DRAWINGS 
     These and other embodiments of the present invention will be readily understood with reference to the following specification and attached drawing wherein: 
     FIG. 1 is a graphical illustration of an MSK modulation signal in the time domain for a short random sequence of bits generated by conventional techniques. 
     FIG. 2 is a graphical illustration of an MSK modulation signal in the time domain for the same short random sequencing bits in FIG. 1 generated by a technique in accordance with one embodiment of the present invention. 
     FIG. 3 is a graphical illustration of the in-phase component of the MSK modulation signal illustrated in FIG. 1. 
     FIG. 4 is a graphical illustration of the in-phase component of the MSK modulation signal illustrated in FIG. 2. 
     FIG. 5 is a graphical illustration of an MSK modulation signal in the frequency domain, generated by conventional techniques. 
     FIG. 6 is similar to FIG. 5, except the MSK modulation signal is generated by a technique in accordance with one embodiment of the invention. 
     FIG. 7 is a block diagram illustrating a technique for generating an MSK modulation signal in accordance with an alternate embodiment of the invention. 
     FIGS. 8a-8g are graphical illustrations of the waveforms in the time domain at very points along the block diagram illustrated in FIG. 7. 
     FIG. 9 is a graphical illustration in the frequency domain comparing a conventional unfiltered MSK signal, an MSK modulation signal utilizing a known technique for reducing sidelobe energy, and an MSK modulation signal generated by a technique in accordance with one embodiment of the present invention. 
     FIG. 10 is a graphical illustration of an MSK modulation signal generated by one technique in accordance with the present invention utilizing a (1-cos) smoothing function compared with an MSK modulation signal generated by a known technique. 
     FIG. 11 is similar to FIG. 10 but for a (1-cos) 2  smoothing function. 
     FIG. 12 is similar to FIG. 10 but for a (1-cos) 1/2   smoothing function. 
     FIG. 13 is a graphical illustration of MSK modulation signals generated by different smoothing functions. 
    
    
     DETAILED DESCRIPTION 
     A conventional minimum shift keying (MSK) modulated signal is generated by linearly increasing or decreasing the phase of a carrier 90°/data bit corresponding to a &#34;0&#34; or &#34;1&#34;. The complex baseband minimum shift keying (MSK) modulation signal is provided in equation (1) below: 
     
         (1) MSK(t)=exp (jπtd(i))/2T, where d(i)=±1 
    
     where the phase term increases or decreases linearly by 90° over each interval T. 
     As shown in FIG. 1, the conventional MSK modulation signal&#39;s phase trajectory is generally triangular over time yielding a frequency spectrum that rolls off roughly as (SIN (f)/f) 2 , when averaged over all sequences. However, as stated above, such conventional MSK modulation signals are known to result in phase derivative discontinuities when successive alternate bits in either the in-phase channel (I) or quadrature channel (Q) have the same value, for example as shown at time T=140 in FIG. 1. Such a situation results in a discontinuity of the first derivative (d.o slashed./dt). As is known in the art, the derivative d.o slashed./dt is the frequency of the modulation signal. Such a discontinuity in the first derivative produces high frequency spectral energy components, manifested by spectral spreading of the MSK modulation signal as shown in FIG. 8. 
     In accordance with a first embodiment of the present invention, to eliminate discontinuities in the first derivative of the MSK modulated signal, a continuous frequency minimum shift keying (CF --  MSK) technique is provided. The CF --  MSK technique smooths the phase discontinuities by increasing or decreasing the phase of the modulation signal in a continuous sinusoidal fashion as set forth in equation 2 below: 
     
         (2) MSK(t)=exp jπtd(i) A sin (πt/T)/2T!, 
    
     where ##EQU3## used to normalize the phase change per chip. The time varying frequency term A sin (πt/T)/2 results in a smooth phase motion with a continuous derivative (i.e. continuous frequency). A CF --  MSK modulation signal is illustrated in FIG. 2. As noted, the phase discontinuities, for example at T=140 have been smoothed out by the sinusoidal function. 
     As mentioned above, the MSK modulation signal can be broken down into two component signals; an in-phase signal (I) and a quadrature signal (Q). The in-phase component signal for both a conventional MSK modulation signal and a CF --  MSK modulation signal in accordance with the present invention is illustrated in FIGS. 3 and 4, respectively. The quadrature component (Q) signals are similar to the in-phase component (I) signals. As shown, the signals are fairly similar except that the sharp signal transitions in FIG. 3 have been eliminated in FIG. 4. 
     FIGS. 5 and 6 illustrate an MSK signal generated by conventional techniques and an MSK signal generated by a technique in accordance with one embodiment of the invention, respectively; both in the frequency domain. As shown, at 50 MHz with 20 samples per data bit--a spectrum of a JTIDS signal with a 5 mbps chip rate from the carrier--the CF --  MSK modulation signal is 20 db lower than the conventional MSK modulation signal. At 21 MHz away from the carrier, the CF --  MSK signal is about 15 db lower in sidelobe energy. As such, in certain applications, the CF --  MSK modulation signal can provide the same level of out-of-band transmitted energy as the conventional MSK signal with less required filtering or even lower out-of-band emissions with the same filtering. For example, in order for the sidelobe energy to be 60 db down at ±14 MHz, the CF --  MSK modulation technique in accordance with the present invention can match this requirement with 5 db filtering where standard MSK filtering techniques require at least 15-20 db of filtering to meet the same value. 
     In an alternate embodiment of the invention, a second derivative of an MSK modulation is taken and convolved with a smoothing function. The resulting signal is then integrated twice to define the smooth phase function .o slashed. (i). The modulation signal in accordance with the alternate embodiment of the invention is generally set forth in equation 3 below: 
     
         (3) S2D.sub.-- MSK(i)=e.sup.j.o slashed.(i) 
    
     The technique for producing a modified MSK modulation signal S2D --  MSK in accordance with equation 3 normally produces discontinuous derivatives beyond the first derivative. However, smoothing of the discontinuity in equation 3 forces all derivatives to be continuous. Thus, the S2D --  MSK modulation technique plus the choice of the smoothing function as set forth below significantly reduces the out-of-band emission without affecting the near spectrum. 
     A technique for generating the S2D --  MSK modulation signal is illustrated in FIG. 7. Initially, as illustrated in block 20, an MSK modulation signal is generated, for example by conventional techniques, by linearly increasing or decreasing the phase of the carrier 90°/data bit corresponding to a binary &#34;0&#34; or &#34;1&#34;. As shown in block 22 and in FIG. 8a, the phase trajectory of the conventional MSK modulation signal has a triangular shape yielding a spectrum that falls off roughly as (SIN (f)/f) 2 . Once the phase modulation signal .o slashed. (t) is generated, a first derivative of the phase modulation signal .o slashed. (t) is taken, as indicated by step 24 and illustrated in FIG. 8b. The first derivative .o slashed.&#39; --  MSK of the modulation signal .o slashed. (t) represents the frequency of the MSK modulation signal. As indicated in the block 26, a second derivative .o slashed.&#34; --  MSK of the modulation signal is taken which, as shown in block 26(FIG. 7) and FIG. 8c, consists of an impulse train. 
     An important aspect of the invention relates to the ability to vary the sidelobe energy in the resulting S2D --  MSK modulation signal as a function of the smoothing function selected. As shown in block 28, a smoothing function is selected, for example s(t)=(1-cos) 2 . The exemplary smoothing function shown is 20 samples long corresponding to a 100 MHz sampling rate. However, other smoothing functions, as discussed below, can be selected to vary the sidelobe energy. Once a smoothing function is selected, the smoothing function s(t) is convolved with the second derivative .o slashed.&#34; --  MSK obtained from block 26 by way of a finite impulse filter (FIR) 30, for example, as disclosed in Alan V. Oppenheim &amp; Schafer, &#34;Digital Signal Processing,&#34; Prentice-Hall, 1975 hereby incorporated by reference. The signal resulting from the convolution of the second derivative signal .o slashed. 11   --  MSK with the smoothing function s(t)=(1-cos) 2  is a signal indicated by the block 32 (FIG. 7) and illustrated in FIG. 8e. The &#34;smooth&#34; second derivative is integrated twice as indicated by the blocks 34, 36, 38 and 40 and illustrated in FIGS. 8f and 8g to produce a signal .o slashed. s  (i) in which the phase change per baud is always exactly 90°. Since the approach operates strictly in the phase domain, the technique produces a constant envelope signal. The final S2D --  MSK modulation signal is then generated by inserting the signal .o slashed. s  into the exponential operator exp(j .o slashed. s  si) to form the in-phase (I) and quadrature (Q) component signals as indicated by the blocks 42 and 44. 
     As mentioned above, other smoothing functions are also contemplated. For example, FIG. 10 illustrates an MSK modulation signal S2D --  MSK utilizing an exemplary smoothing function s(t)=(1-cos). FIGS. 10-12 illustrate the modulation signals S2D-MSK in accordance with the present invention for smoothing functions s(t)=(1-cos); s(t)=(1-cos) 2  and s(t)=(1-cos) 1/2   respectively, compared with an MSK modulation signal generated by a known technique for reducing sidelobe energy. As shown, at ±20 MHz from the carrier, the sidelobe energy of the MSK modulation signals S2D --  MSK is relatively less than the sidelobe energy generated by the known technique. As shown best in FIG. 13, the sidelobe energy for the modulation signals S2D --  MSK with different smoothing functions also varies. For example, at 50 MHz from the carrier frequency, the spectral roll off varies from approximately -140 dBc to about -200 dBc. Thus, by storing various smoothing functions, the spectral roll off and hence the sidelobe energy can be controlled as a function of the selected smoothing function, which can be stored in a lookup table. 
     Obviously, many modifications and variations of the present invention are possible in light of the above teachings. Thus, it is to be understood that, within the scope of the appended claims, the invention may be practiced otherwise than as specifically described above.