Abstract:
A receiver of baseband signals from a communications line characterized by baseline wander, including a pre-decoding section, which receives and samples the signals and subtracts each sample from a preceding sample so as to generate corrected data, and an equalization section, which receives the corrected data and generates equalized output data representative of data input to the line and generally free of the baseline wander. The receiver preferably includes an A/D converter, which digitizes the signals either before or after pre-decoding.

Description:
FIELD OF THE INVENTION 
     The present invention relates generally to digital signal processing, and specifically to digital receivers for AC-coupled lines. 
     BACKGROUND OF THE INVENTION 
     Local-area networks (LANs) or communication devices transmitting and receiving digital signals commonly operate on standards such as Ethernet 10BASE-T or 100BASE-TX. The 100BASE-TX Ethernet standard enables communication at 100 Mb/s on unshielded twisted pair (UTP) copper wire by using MLT-3 encoding. MLT-3 encoding transmits “1”s as ordered level changes between 3 levels {1, 0, −1}, whereas “0”s are transmitted as the same level as the previous symbol. Thus the signal 1111111 could be encoded as {0, 1, 0, −1, 0, 1, 0, −1}, and the signal 1111011100 could be encoded as {0, 1, 0, −1, −1, 0, 1, 0, 0, 0}. In principle, other forms of ordered level change encoding can also be used. For example, instead of 3 ordered level changes, signals could also be encoded with 5 ordered level changes {2, 1, 0, −1, −2}. 
     One of the advantages of ordered level change encoding is that the high frequency components of the signal are reduced. For MLT-3 encoding with signals clocked at a standard rate of 125 MHz (8 ns per symbol), the signal frequency varies from 0, for a run of “0”s, to a maximum of 31.25 MHz (125/4) for a run of “1”s. (125 MHz is a nominal frequency, and in practice the frequency will vary slightly from the nominal.) The relatively low signal frequency is advantageous in reducing electromagnetic interference (EMI) and relaxing frequency-related demands on signal processing equipment and wiring. However, MLT-3 encoding creates inherent problems for receivers, particularly when the receivers are at the end of long runs (of the order of 100 m) of cable, as described hereinbelow. 
     MLT-3 signals are transmitted and received via transformers, so that there is no path for DC between transmitter and receiver. If a continuous string of “0”s is transmitted, then there may be an effective DC level in the transmitted signal, which needs to be detected by the receiver. At the receiver, the signal is detected by digitizing and comparing the received signal to the receiver&#39;s baseline. In order to correctly detect DC levels, the receiver&#39;s baseline must be constantly adjusted for baseline wander (BLW)—since BLW or the inaccurate correction thereof causes errors in the recovered signal. 
     The incoming signal is sampled and digitized by an A/D converter, preferably operating at the minimum theoretical sample rate for the A/D converter, equal to the clock rate of the signal, i.e., the nominal 125 MHz. In order for the A/D converter to operate efficiently, the receiver has to recover the exact clock timing, both in frequency and in phase, from the received signal. 
     In a paper by Mueller and Muller, “Timing recovery in digital synchronous data receivers,” IEEE Transactions on Communications, pp 516-531, Vol. 24, May 1976, which is herein incorporated by reference, the authors propose a timing recovery algorithm. The paper is accepted in the art as the basis for timing recovery algorithms, and relies on selecting a timing function that is zero at an assumed best sampling point. The phase of the sampling point is then adjusted until its phase is zero. 
     In a paper by Fertner and Solve, “Symbol-rate timing recovery comprising the optimum signal-to-noise ratio in a digital subscriber loop,” IEEE Transactions on Communications, pp 925-936, Vol. 45, August 1997, which is herein incorporated by reference, the authors investigate a recovery algorithm that is based on the correlation between a mean-square error from a decision feedback equalizer and an arriving sample signal. The authors also point out practical complications involved in the relatively conceptually straightforward derivation of Mueller and Muller. 
     FIG. 1 is a graph showing the typical received shape of an 8 ns positive pulse after transmission along different lengths of unshielded twisted pair category  5  (UTP cat- 5 ) cables. The pulse, comprising a sharp leading edge and a less sharp trailing edge, drops in height exponentially, and increases in width with increasing cable length. Consequently, for cable lengths over 100 m, it becomes increasingly difficult to recover the clock and distinguish one pulse from the next. 
     FIG. 2 shows a composite received signal 11 for a cable 130 m long, given an input signal  13  of 1, 1, 1, 1, 0, −1, 0, 1, wherein 1 corresponds to a positive pulse and −1 corresponds to a negative pulse. The circles on composite graph  11  correspond to measured signals spaced 8 ns apart. This graph illustrates the difficulty of recovering the clock and the input signal values, since the measured values are not simply related to the input signal of 1, 1, 1, 1, 0, −1, 0, 1. 
     FIG. 3 is a block diagram of a receiver  20  used to detect 100BASE-TX signals of the type shown in FIG. 2, as is at present known in the art. Signals from a magnetics (transformer) stage are input to an automatic gain control (AGC) amplifier  14 , and transferred to an analog summer  18 , wherein a BLW correction is added. The result is transferred to an A/D converter  21 . The A/D converter generates corresponding digital signals, sampled according to an input clock signal from a PLL  40  and phase multiplexer  42 , and the digitized signals are transferred to a digital signal processing (DSP) core  48 . The clock signal is synchronized in frequency and phase with the incoming input signal, in order to minimize conversion errors in the A/D converter. 
     DSP core  48  comprises a forward equalization (FEQ) module  26 , an adder  28 , a decision (DEC) module  30 , and a decision feedback equalizer (DFE) module  32 , which together act to supply data to a baseline wander correction module  24 . BLW correction module  24  supplies the aforementioned (analog) BLW correction signal to summer  18 . Typically, the magnetics stage has a non-linear inductance, and acts as a high pass filter, and BLW module  24  comprises a matching low pass filter whose frequency response is adjustable. The characteristics of the low pass filter are pre-adjusted to minimize BLW. The high pass filter characteristics of the magnetics stage, however, depend on the DC current flowing in the magnetics stage, so that the characteristics are not fixed and are difficult to predict. 
     DSP core  48  also comprises a DSP control  36  and a timing control  38 . On the basis of signals output by decision module  30 , DSP control  36  supplies data to timing control  38 . Timing control  38  controls the frequency and phase of the clock signal supplied by multiolexer  42 , for example, according to the aforementioned method of Mueller and Muller. Core  48  transfers the equalized, BLW-corrected signals in MLT-3 format to module  46 , wherein the signals are processed further for transmission in binary format, preferably in a non-return-to-zero (NRZ) format. 
     Other existing receivers use analog equalizers, such as high pass filters; these equalizers inherently enhance the noise at the same time as they enhance the high-frequency gain. Errors in the assumed parameter values of the equalizers lead to an error in reconstructing the BLW. Furthermore, any decision error leads to symbol error and inaccurate BLW correction for a relatively long time period. 
     In order to overcome the inherent limitations of poor transmission of low frequency signals through the input transformers, existing receivers use complicated adaptive algorithms to reconstruct the transmitted DC level. Existing receivers continuously monitor the signal baseline to correct BLW. 
     SUMMARY OF THE INVENTION 
     It is an object of the present invention to provide an improved receiver for high frequency digital signals. 
     It is a further object of some aspects of the present invention to provide methods and devices for substantially eliminating the effects of baseline wander in a receiver. 
     It Is a further object of some aspects of the present invention to provide improved methods and devices for synchronizing a receiver clock with an input signal clock rate. 
     It is a yet further object of some aspects of the present invention to provide improved methods and devices for equalizing received signals. 
     In preferred embodiments of the present invention, a receiver comprises an A/D converter with a variable reference, a pre-decoding section, a digital equalization section, and an output section. The A/D converter preferably accepts 100BASE-T signals, and the output section preferably outputs the signals in NRZ format. Signals input to the receiver are transferred directly to the A/D converter, with substantially no intervening signal adjustment for baseline wander, unlike receivers at present known in the art. The necessity for additional compensation for baseline wander is substantially eliminated by the pre-decoding section, wherein each signal sampled and digitized by the A/D converter is subtracted from a preceding sample, thus substantially eliminating the effects of baseline wander (BLW). 
     In preferred embodiments of the present invention, the A/D converter is placed before the pre-decoding section. Alternatively, the pre-decoding section is placed before the A/D converter. 
     In some preferred embodiments of the present invention, the equalization section has a unique pipeline architecture, enabling it to operate at substantially faster clock rates, and with substantially fewer components, compared to equalizers known in the art. The equalization section comprises both forward equalization and decision feedback equalization stages on a common pipeline, with multiplicative coefficients determined using an adaptive process, preferably a least mean squares adaptation. Preferably, clock recovery from the incoming signal is performed by measuring differences between two or more of the coefficients evaluated in the equalization section, using the differences to give substantially better clock recovery for weaker signals than methods at present known in the art. Most preferably, differences are measured between one coefficient in the decision feedback equalization stage, and one coefficient in the forward equalization stage. 
     There is therefore provided, in accordance with a preferred embodiment of the present invention, a receiver of baseband signals from a communications line characterized by baseline wander, including: 
     a pre-decoding section, which receives and samples the signals and subtracts each sample from a preceding sample so as to generate corrected data; and 
     an equalization section, which receives the corrected data and generates equalized output data representative of data input to the line and generally free of the baseline wander. 
     Preferably, the receiver includes an A/D converter which digitizes the signals and transmits the digitized signals to the pre-decoding section. Alternatively, the A/D converter digitizes the corrected data and transmits the digitized corrected data to the equalization section. 
     Preferably, the equalization section includes a pipeline, including one or more delay stages and one or more respective adders, and the pre-decoding section includes an input to the pipeline. 
     Preferably, the baseband signals are encoded in an ordered level change format, most preferably an MLT-3 format. 
     Preferably, the equalization section includes a decision module which compares the equalized data to one or more predetermined thresholds and responsive thereto outputs decision data corresponding to a level of the input data. In a preferred embodiment, the decision module outputs an error signal, indicative of a deviation of the equalized output data relative to the level of the input data, which error signal is fed back to an input of the equalization section. Preferably, the equalization section generates, responsive to the error signal, one or more forward equalization coefficients, which multiply the corrected data, and one or more decision feedback equalization coefficients, which multiply the decision data, and the multiplied corrected data and decision data are summed to generate the equalized data. Further preferably, the receiver includes a clock generator, which provides a timing signal to control the sampling of the A/D converter, wherein the generator adjusts the timing signal responsive to one or more of the coefficients. 
     There is also provided, in accordance with a preferred embodiment of the present invention, a receiver of baseband signals from a communications line, including: 
     an A/D converter, which samples and digitizes the signals to generate digitized data; 
     an equalization section, which receives the digitized data and generates equalized output data representative of data input to the line; 
     a decision module, which compares the equalized data to one or more predetermined thresholds and responsive thereto outputs decision data indicative of a level of the data input to the line; and 
     a clock generator which generates a variable clock signal responsive to the decision data, which clock signal is used to time the sampling of the A/ID converter. 
     Preferably, a phase of the clock generator is varied responsive to the decision data. Most preferably, the clock generator provides a plurality of clock signals having different, respective phases, such that the phase of the clock generator is varied by selecting one of the plurality of signals responsive to the decision data. 
     Additionally or alternatively, a frequency of the clock generator is varied responsive to the decision data. 
     Preferably, the clock signal is generated responsive to an error signal indicative of a deviation of the equalized output data relative to the data input to the line. 
     In a preferred embodiment, the equalization section includes a processing pipeline, which generates, responsive to the error signal, one or more equalization coefficients, including one or more forward equalization coefficients which multiply the digitized data, and one or more decision feedback equalization coefficients, which multiply the decision data, and the clock signal is generated responsive to one or more of the coefficients. Most preferably, the clock generator generates the clock signal responsive to a precursor coefficient of the one or more forward equalization coefficients and a most significant one of the one or more decision feedback equalization coefficients. 
     Further preferably, the clock generator generates the clock signal responsive to an integration of the at least one of the coefficients over a predetermined number of clock cycles, wherein the clock signal is varied responsive to a primary difference between the integration and the at least one of the coefficients. Alternatively or additionally, the clock signal is varied responsive to a secondary difference corresponding to a variation over time in the primary difference. 
     In a preferred embodiment, the clock generator generates the clock signal responsive to a difference between one of the forward equalization coefficients and one of the decision feedback equalization coefficients. Preferably, the clock generator generates a frequency offset of the clock signal responsive to an integration over a predetermined number of clock cycles of the difference between one of the forward equalization coefficients and one of the decision feedback equalization coefficients. Alternatively or additionally, the clock generator generates a phase change of the clock signal responsive to at least one integration of the difference between one of the forward equalization coefficients and one of the decision feedback equalization coefficients. 
     There is further provided, in accordance with a preferred embodiment of the present invention, a receiver of baseband signals from a communications line, including: 
     an A/D converter, which samples and digitizes the signals to generate digitized data; 
     an equalization section, including a pipeline which receives the digitized data and generates equalized output data representative of data input to the line, the pipeline including a plurality of multipliers, which multiply data input thereto by respective multiplication coefficients, and a plurality of adders, which receive and sum the multiplied data; and 
     a decision module, which compares the equalized output data to one or more predetermined thresholds so as to generate decision data indicative of a level of the input data, which decision data are input to the pipeline together with the digitized data. 
     Preferably, the pipeline includes a plurality of delay registers, intermediate the adders, which transfer the data from one of the adders to the next in the pipeline. 
     In a preferred embodiment, the decision module generates an error signal responsive to a deviation of the equalized data relative to the decision data, and the multipliers multiply the digitized data and the decision data by respective coefficients generated by the equalization section responsive to the error signal. 
     Preferably, one or more of the coefficients are generated by multiplying the error signal by the digitized data or, alternatively or additionally, by multiplying the error signal by the decision data. 
     Preferably, each of at least some of the adders in the pipeline receives and sums a respective one of the multiplied digitized data and a corresponding one of the multiplied decision data, wherein at least one of the at least some of the adders receives and sums the respective multiplied digitized data and multiplied decision data together with an output of a preceding one of the adders in the pipeline. 
     In a preferred embodiment, the pipeline includes a pre-decoding section, which subtracts each of the input data from a preceding one of the data so as to substantially eliminate baseline wander from the signals. 
     There is additionally provided, in accordance with a preferred embodiment of the present invention, a method for processing baseband signals from a communications line characterized by baseline wander, including: 
     receiving and sampling the signals and subtracting each sample from a preceding sample, in a pre-decoding section, so as to generate corrected data; and 
     receiving the corrected data, in an equalization section, and generating equalized output data therefrom representative of data input to the line and generally free of the baseline wander. 
     Preferably, the method includes digitizing the signals in an A/D converter and transmitting the digitized signals to the pre-decoding section or, alternatively, digitizing the corrected data in an A/D converter and transmitting the digitized corrected data to the equalization section. 
     Preferably, generating equalized output data includes passing the data through a pipeline, including one or more delay stages and one or more respective adders, and subtracting each sample includes inverting each sample and inputting the inverted sample to the pipeline. 
     Preferably, receiving the signals includes receiving signals encoded in an ordered level change format, most preferably an MLT-3 format. 
     Preferably, the method includes comparing the equalized data to one or more predetermined thresholds and responsive thereto outputting decision data corresponding to a level of the input data, wherein comparing the data preferably includes outputting an error signal, indicative of a deviation of the equalized output data relative to the level of the input data, and wherein equalizing the data includes processing the data responsive to the error signal. 
     In a preferred embodiment, equalizing the data includes generating, responsive to the error signal, one or more forward equalization coefficients, which multiply the corrected data, and one or more decision feedback equalization coefficients, which multiply the decision data, and summing the multiplied corrected data and decision data. Preferably, the method further includes generating a clock signal to time the sampling of the signals, wherein the clock signal is adjusted responsive to one or more of the coefficients. 
     There is moreover provided, in accordance with a preferred embodiment of the present invention, a method of processing baseband signals received from a communications line, including: 
     sampling and digitizing the signals to generate digitized data; 
     determining one or more equalization coefficients responsive to a level of the digitized data; 
     equalizing the digitized data to generate equalized output data representative of data input to the line by multiplying the digitized data by the one or more equalization coefficients; and 
     generating a variable clock signal responsive to at least one of the one or more equalization coefficients, which clock signal is used to time the sampling. 
     Preferably, generating the clock signal includes varying a phase of the clock signal responsive to the decision data, most preferably by providing a plurality of clock signals having different, respective phases, and selecting one of the plurality of signals responsive to the decision data. 
     Alternatively or additionally, generating the clock signal includes varying a frequency of the clock signal responsive to the decision data. 
     In a preferred embodiment, determining the one or more coefficients includes generating an error signal indicative of a deviation of the equalized output data relative to the data input to the line and determining one or more of the coefficients responsive to the error signal. Preferably, determining the one or more equalization coefficients includes determining one or more forward equalization coefficients and one or more decision feedback equalization coefficients, and equalizing the data includes multiplying the digitized data by the one or more forward equalization coefficients and multiplying the decision data by the one or more decision feedback equalization coefficients and adding the multiplied data together in a pipeline, and generating the clock signal is performed responsive to one or more of the coefficients. Most preferably, generating the clock signal includes generating a signal responsive to a precursor coefficient of the one or more forward equalization coefficients and a most significant one of the one or more decision feedback equalization coefficients. 
     In a preferred embodiment, generating the clock signal includes integrating at least one of the coefficients over a predetermined number of clock cycles to generate an integrated output and varying the clock signal responsive to the integrated output. Preferably, varying the clock signal includes determining a primary difference between the integrated output and the at least one of the coefficients and varying the clock signal responsive to the primary difference. Additionally or alternatively, varying the clock signal includes determining a secondary difference corresponding to a variation over time in the primary difference and varying the clock signal responsive to the secondary difference. 
     In a preferred embodiment, generating the clock signal includes varying the clock signal responsive to a difference between one of the forward equalization coefficients and one of the decision feedback equalization coefficients. Preferably, varying the clock signal includes generating a frequency offset of the clock signal responsive to an integration over a predetermined number of clock cycles of the difference between the one of the forward equalization coefficients and the one of the decision feedback equalization coefficients. Alternatively or additionally, varying the clock signal includes generating a phase change of the clock signal responsive to at least one integration of the difference between the one of the forward equalization coefficients and the one of the decision feedback equalization coefficients. 
     There is further provided, in accordance with a preferred embodiment of the present invention, a method of processing baseband signals received from a communications line, including: 
     sampling and digitizing the signals to generate digitized data; 
     equalizing the digitized data by processing the data in a pipeline to generate equalized output data representative of data input to the communications line, which processing includes: 
     multiplying data input to the pipeline by a plurality of respective multiplication coefficients; and 
     summing the multiplied data together in the pipeline; 
     comparing the equalized output data to one or more predetermined thresholds so as to generate decision data indicative of a level of the input data; and 
     inputting the decision data to the pipeline together with the digitized data. 
     Preferably, multiplying and summing the data include multiplying and summing data in a plurality of pipeline stages, and equalizing the data includes delaying the data in the pipeline between one stage and the next. 
     In a preferred embodiment, comparing the equalized data comprises generating an error signal responsive to a deviation of the equalized data relative to the decision data, and multiplying the digitized data and the decision data includes multiplying the data by coefficients generated responsive to the error signal. Preferably, multiplying the data includes multiplying the data by coefficients generated by multiplying the error signal by the digitized data. Additionally or alternatively, multiplying the data includes multiplying the data by coefficients generated by multiplying the error signal by the decision data. 
     Preferably, summing the data includes summing a respective one of the multiplied digitized data and a corresponding one of the multiplied decision data at one or more stages in the pipeline. Preferably, summing the data at the one or more stages includes summing the multiplied digitized data and the multiplied decision data together with an output of a preceding stage in the pipeline. 
     In a preferred embodiment, the method includes substantially eliminating baseline wander from the signals by subtracting each of the data input to the pipeline from a preceding one of the data input to the pipeline. 
     The present invention will be more fully understood from the following detailed description of the preferred embodiments thereof, taken together with the drawings in which: 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a graphical representation of the impulse response of a line to pulses conveyed over cables of different lengths; 
     FIG. 2 is a graphical representation of a composite pulse signal conveyed over a cable of length 130 meters; 
     FIG. 3 is a block diagram of a receiver as is at present known in the art: 
     FIG. 4A is a block diagram of a receiver for MLT-3 signals in accordance with a preferred embodiment of the present invention; 
     FIG. 4B is a block diagram of a receiver for MLT-3 signals in accordance with an alternative preferred embodiment of the present invention; 
     FIG. 5 is a block diagram of a forward equalizer and a decision feedback equalizer, for use in the receiver of FIG. 4A or the receiver of FIG. 4B, in accordance with a preferred embodiment of the present invention; 
     FIG. 6 is a block diagram of a combined forward equalizer and decision feedback equalizer, for use in the receiver of FIG. 4A or the receiver of FIG. 4B, in accordance with a preferred embodiment of the present invention; 
     FIG. 7 is a block diagram of an alternative combined forward equalizer and decision feedback equalizer, for use in the receiver of FIG. 4A or the receiver of FIG. 4B, in accordance with a preferred embodiment of the present invention; 
     FIG. 8 is a block diagram of a timing controller, for use in the receiver of FIG. 4A or the receiver of FIG. 4B, in accordance with a preferred embodiment of the present invention; 
     FIG. 9 is a block diagram of an alternative timing controller, for use in the receiver of FIG. 4A or the receiver of FIG. 4B, in accordance with a preferred embodiment of the present invention; 
     FIG. 10 is a block diagram of a section of the equalizers of FIG. 6, showing the generation of coefficients and the operation of one tap of the equalizers, in accordance with a preferred embodiment of the present invention; and 
     FIG. 11 is a block diagram of a decision module, for use in the receiver of FIG. 4A or the receiver of FIG. 4B, in accordance with a preferred embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     Reference is now made to FIG. 4A, which is a block diagram of a receiver  60 , in accordance with a preferred embodiment of the present invention. Receiver  60  receives 100BASE-T signals in MLT-3 format from a magnetics stage (not shown) on an input line  62  to an A/D converter  70 , without intervening input for baseline wander (BLW) correction, wherein the analog signal is converted to digital signals output on a plurality of lines  92 . A/D converter  70  preferably has an 8-level selectable dynamic range, which is selected according to signals from a digital signal processing (DSP) controller module  86 . Preferably the dynamic range of A/D converter  70  is changed by adjusting a reference voltage of the converter. Receiver  60  is preferably implemented in a single custom integrated circuit chip, although discrete components or a combination of discrete and custom or semi-custom components may similarly be used. 
     Preferably A/D converter  70  comprises a six-bit converter and samples the incoming signals according to a clock signal derived from a phase-locked-loop clock  76 . Clock  76  supplies a plurality of clock signals to a multiplexer  74 , preferably at least sixteen different clock signals at a frequency of about 125 MHz, each signal having a respective, different phase. Multiplexer  74  outputs one of the clock signals, chosen according to data supplied to multiplexer  74  from a timing controller  84 , to A/D converter  70 . The generation of controlling signals to timing controller  84  is described in more detail hereinbelow. 
     Signals from A/D converter  70  are sequentially transferred to a pre-decoding section  91 , which takes the place of BLW correction  24  shown in FIG.  3  and typically used in receivers known in the art. Pre-decoding section  91  comprises a delay register  96  and an adder  98  with an inverting input. Most preferably delay register  96  operates at the incoming signal baud rate. Delay register  96  delays incoming signals by one clock period, so that a signal output from adder  98  is the result of subtracting each signal sample from an immediately preceding signal sample. Thus any “DC” level signal, such as a run of “1”s, or a run of “−1”s, will be output as a zero level from adder  98 , substantially eliminating the baseline wander effect that is present in existing receivers. 
     Signals output from adder  98  are input to an equalization section  93 . Section  93  comprises an equalization module  90 , including forward equalization (FEQ) and decision feedback equalization (DFE) functions, and a decision module  88 . Module  90  equalizes the signals received from adder  98 , and also supplies an input to timing controller  84 , as noted above. Module  90  furthermore acts as a whitening filter, thereby flattening the spectrum and so improving the convergence of the equalization. (In the book “Digital Communication,” by Edward Lee and David Messerschmitt, Kluwer Academic Publishers, which is herein incorporated by reference, the authors describe the advantages of using whitening filters in adaptive equalizers.) The equalized signals are input to decision module  88 , which determines whether or not there has been a transition in the MLT-3 signal (indicating a “1,” as described hereinabove). An output from module  88  is input to module  90 , for use in the decision feedback equalization. The structure and operation of module  90  and module  88  are described in greater detail hereinbelow. Signals from module  88  are decoded into a standard binary format, preferably non-return-to-zero (NRZ) format, in a NRZ module  104 , and transmitted for further processing. Module  104 , and respective input lines  102  and output lines therefrom, make up an output section  95 . 
     Receiver  60  also comprises a digital signal processor module  86 , which controls timing controller  84 , decision module  88 , module  90  and delay module  96 , and supplies signals to A/D converter  70  to determine in which input voltage range the converter operates. 
     FIG. 4B is a block diagram of a receiver  60 ′, in accordance with an alternative preferred embodiment of the present invention. Apart from the differences described hereinbelow, the operation of receiver  60 ′ is generally similar to that of receiver  60  (FIG.  4 A), whereby elements indicated by the same reference numerals in both receivers  60  and  60 ′ are generally identical in construction and in operation. In receiver  60 ′ the positions of pre-decoding section  91  and A/D converter  70  are reversed compared to their positions in receiver  60 . Signals in MLT-3 format are received by an analog delay line  97  of pre-decoding section  91 , which delays the signals by a single clock period, and are then subtracted from corresponding undelayed signals by a summer  99 . A signal from summer  99  is input to A/D converter  70 . (It will be appreciated that the dynamic range required of A/D converter  70  in receiver  60 ′ is consequently less than the dynamic range of A/D converter in receiver  60 , for similar signals.) Signals from A/D converter  70  are transferred to equalization section  93 , which operates on the signals substantially as described hereinabove for receiver  60 . 
     FIG. 5 is a block diagram of equalization module  90 , in accordance with a preferred embodiment of the present invention. Module  90  comprises a forward equalization (FEQ) section  112 , and a decision feedback equalization (DFE) section  114 . Section  112  comprises a plurality of FEQ coefficient blocks  118 , having adaptively variable coefficients “Coeff f1”, “Coeff f2”, . . . , through “Coeff f7”; a plurality of FEQ coefficient multipliers  120 ; a plurality of single clock delays  122 ; and a plurality of adders  124 . It will be observed that section  112  operates as a forward equalizer for input signals “X i ” received from adder  98 . Section  114  likewise comprises a plurality of DFE coefficient blocks  128 , having adaptively variable coefficients “Coeff d1”, “Coeff d2”, . . . , through “Coeff d7”; a plurality of DFE coefficient multipliers  130 ; a plurality of single clock delays  132 ; and a plurality of adders  134 . Section  114  thus operates as a decision feedback equalizer for signals “Dec i ” output by decision module  88 . (The generation of the FEQ coefficients and of the DFE coefficients is described in detail hereinbelow.) The outputs of section  112  and section  114  are summed by an adder  136 , and the result transferred to decision module  88 . 
     FIG. 6 is a block diagram of a forward equalization and decision feedback module  90 ′, in accordance with an alternative preferred embodiment of the present invention. This embodiment is functionally similar to the embodiment shown in FIG. 5, but reduces substantially the number of adders and delay register elements that need to be used. In FIG. 6, module  90 ′ comprises coefficient blocks  118  and multipliers  120  in an FEQ section  158 , and coefficient blocks  128  and multipliers  130  in a DFE section  160 . Module  90 ′ further comprises a plurality of adders  152 , and a plurality of single clock delays  154 , in the form of a single series pipeline. In distinction from the operation of module  90 , outputs of corresponding multipliers  120  and  130  are added by their respective corresponding adder  152 , and the result transferred via the respective corresponding clock delay  154  to the next adder  152  in the pipeline. The process continues for the plurality of adders  152 , until the final equalized signal is output from a final adder  156  to decision module  88 . Furthermore, the pipeline architecture of module  90 ′ means that a very fast clock rate may be used in the module. 
     FIG. 7 is a block diagram of a forward equalization and decision feedback module  170 , in accordance with an alternative preferred embodiment of the present invention. Module  170  substantially performs the functions ascribed hereinabove to module  90 , as shown in FIG.  4 A and FIG. 4B, using a combined pipeline architecture such as that shown in FIG. 6, but with relatively fewer FEQ stages. Module  170  comprises a forward equalization section  172 , and a decision feedback equalization section  174 . Section  172  comprises a most significant FEQ coefficient block  178  and a precursor FEQ coefficient block  179 , respectively having adaptively variable coefficients “Coeff B1” and “Coeff B2”, and further comprises a plurality of, preferably two, FEQ coefficient multipliers  180 . Section  174  comprises a plurality, preferably  11 , of DFE coefficient blocks  182 , respectively having adaptively variable coefficients “Coeff d1” through “Coeff d11” (not all shown in FIG.  7 ), a most significant DFE coefficient block  183  having coefficient “Coeff d12”, and a plurality of respective DFE coefficient multioliers  184 . Module  170  also comprises a plurality of single clock delay blocks  186 , a plurality of adders  188 , and multiple-input adders  190 ,  192 , and  194 . A resultant equalized signal is output from adder  194 . 
     In addition to equalizing the signals X i , module  170  also provides timing information to timing controller  84  of FIG. 4A or FIG.  4 B. The timing information is provided by generating a phase error signal from the difference between most significant DFE coefficient  183  and precursor FEQ coefficient  179 . The operation of timing controller  84  is described in greater detail hereinbelow. 
     FIG. 8 is a block diagram of timing controller  84  of FIG. 4A or FIG. 4B, in accordance with a preferred embodiment of the present invention. Signals corresponding to the most significant DFE coefficient “Coeff d12,” from module  170 , enter a first adder  282 , whose output is transferred to a second adder  284 . The signals from adder  284  are fed back to adder  282 , so that adders  282  and  284  act together as an integrator. A counter  288  receives clock signals from DSP controller  86  (shown in FIG.  4 A and FIG.  4 B), and acts as a modulo  8  counter. After eight cycles counter  288  outputs to adder  284  to stop the integration, and to reset adder  284  to zero. The integrated “Coeff d12” output from adder  284  is fed forward via a closed switch  286  to a timing filter  290 . Switch  286  is normally open, and is closed by a signal from counter  288 . 
     Timing filter  290  also receives, from module  170 , signals corresponding with the precursor FEQ coefficient “Coeff B2.” Filter  290  calculates and stores a primary tri-level difference (1, 0, or −1) between precursor FEQ coefficient “Coeff B2” and the integrated “Coeff d12” output. Filter  290  then evaluates a secondary difference between the present tri-level difference and a previously stored tri-level difference. Using the values of the secondary difference and the present and previous primary differences, filter  290  outputs a clock phase change signal and a frequency offset signal to multiplexer  74  (shown in FIG.  4 A and FIG.  4 B). The phase change and the frequency offset are chosen so as to iteratively minimize the primary and secondary differences, according to the condition that at each iteration the phase change is zero or an increment or a decrement of {fraction (1/16)} of a clock cycle. 
     The phase change signal from filter  290  is generated according to a table  292  included within the filter, whose characteristics are shown hereinbelow, wherein a phase change of +1 corresponds to a signal to increment the phase of the clock signal by {fraction (1/16)} of a cycle, a phase change of −1 corresponds to a signal to decrement the phase of the clock signal by {fraction (1/16)} of a cycle, and a phase change of 0 corresponds to no change in the phase of the clock signal: 
     
       
         
               
               
               
             
               
               
               
             
               
               
             
           
               
                   
               
               
                 Present 
                 Previous 
                 Phase 
               
               
                 Difference 
                 Difference 
                 Change 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 0 
                 +1 
                 −1 
               
               
                 0 
                 −1 
                 +1 
               
               
                 +1 
                 0 
                 +1 
               
               
                 −1 
                 0 
                 −1 
               
             
          
           
               
                 No combination above 
                 0 
               
               
                   
               
             
          
         
       
     
     The frequency offset signal from filter  290  is thus generated by iteratively solving an equation df=df+k1*dp, wherein df is a preliminary frequency offset, dp is the primary tri-level difference between precursor FEQ coefficient “Coeff B2” and the integrated “Coeff d12” output, and k1 is a predetermined constant. 
     FIG. 9 is a block diagram of a timing controller  384 , in accordance with a preferred embodiment of the present invention. Timing controller  384  may be used in place of timing controller  84 , in receiver  60 . Signals corresponding to the most significant DFE coefficient “Coeff d12” and to precursor FEQ coefficient “Coeff B2” from module  170 , enter a first adder  382 , which outputs a preliminary phase evaluation dp at block  386 . dp is determined according to the equation: dp=B2−d12−DPbase, wherein DPbase is a constant offset value, preferably equal to 0.0625. 
     Signals from block  386  transfer to a second-order loop  387  (which performs the function of filter  290  in FIG.  8 ). Loop  387  includes an adder  388 , which together with a shift-right block  390  and an initial frequency determining block  398 , iteratively calculates a preliminary frequency offset df. The offset is calculated according to the equation: df=df+k1*dp, wherein K1 is a coefficient that is determined according to the shift applied by block  390 . Preferably, block  390  shifts right by 13 places. 
     Adder  388  accumulatively adds the value of df and the value of dp, to generate a frequency offset signal, which is further transferred, via a shift-right block  392 , to an adder  394  and a phase integrator block  396 . Preferably, block  392  shifts right by 7 places. The result of the integration is output to a difference storage register  400 . The current and previous values from register  400  (wherein the previous value is generated using a delay register  402 ) are input to a table  292 , whose characteristics are substantially similar to those described hereinabove for timing controller  84 . The output from the table determines the phase change signal output from timing controller  384  to multiplexer  74  (FIGS.  4 A and  4 B). 
     The use of one FEQ coefficient and one DFE coefficient in a system substantially as described hereinabove for controller  84  or controller  384 , in order to correct both the frequency and the phase of the clock signal, leads to substantially better recovery and stability of the clock signal compared to systems at present known in the art. 
     FIG. 10 is a block diagram showing a detail of module  90 ′ of FIG. 6, in accordance with a preferred embodiment of the present invention, showing the internal operation of a section  200  of the module and the generation of FEQ coefficients and DFE coefficients therein. Module  90 ′ may be considered to be constructed as a plurality of sections substantially similar to section.  200 , connected sequentially. Section  200  corresponds to a third tap  201  of section  158  and to a third tap  203  of section  160  of FIG. 6. A subsection  202  of section  200  receives a signal X n  from a previous tap, which enters a single clock delay block  204 . The signal is then transferred to a next tap along section  158  and is also input to a multiplier  206 , wherein it is multiplied by an error signal derived from decision module  88  of FIG.  4 . The generation of the error signal is described hereinbelow. 
     Multiplier  206  transfers its output to a shifter  208 , which divides the output of the multiplier by a predetermined power of 2 and outputs the result to a first input of an adder  210 . Adder  210  outputs its result to a single clock delay block  212 , which outputs the respective FEQ coefficient. The FEQ coefficient is fed back to a second input of adder  210 , which thus acts as a integrator, and is fed forward to a multiplier  214 , to which signal X i  is also input. The multiplicand of multiplier  214  is transferred to a first input of adder  152 . It will thus be understood that as long as the error signal input to multiplier  206  is non-zero, the FEQ coefficient will gradually change, so as to improve the equalization of the signal. When the error signal is zero, the FEQ coefficient will stabilize at a substantially optimal value. 
     Section  204  operates on signals Decn in substantially the same way as described hereinabove for section  202 , outputting its result to a second input of adder  152 . It will be appreciated that in section  204  signals Decn have values 1, 0, or −1, so that multiplier  226  simply acts as a selector for its incoming error signal, outputting either the error signal itself, or its complement, or zero. 
     Adder  152  receives a third equalized input Eq n−1  from a previous tap  205 , and the output of adder  152  is transferred to single clock delay  154 . The output of single clock delay  154  is an equalized output Eqn of section  200 , and output E qn  is transferred forward to a following section, or alternatively, as the final equalized signal of module  90 ′. While the description hereinabove for the operation of section  200  applies specifically to module  90 ′, it will be appreciated that the operation of a section  209  of module  170  (shown in FIG. 7) will be substantially the same as the operation of section  200 . 
     FIG. 11 is a block diagram of decision module  88  of FIG. 4A or FIG. 4B, in accordance with a preferred embodiment of the present invention. Decision module  88  decides which of three levels, 1, 0, or −1, an incoming equalized signal represents, by comparing the incoming signal to a first reference level of +½, and to a second reference level of −½. Module  88  furthermore generates the error signal that is input, as described hereinabove, to multiplier  206  and selector  226  of section  200 , and which is substantially dependent on the difference between the incoming equalized signal and an output signal “Dec”. 
     Equalized signals from FEQ/DFE block  90  are input to comparators  254  and  256 , and are also input to a junction  266 . Comparator  254  compares the signal to the reference +½ level, and comparator  256  compares the signal to the reference −½ level. The comparison is exemplified by a schematically illustrated signal  252 . The outputs of the respective comparators are output to a logic module  258 , which determines the level, 1, 0, or −1, to which to set the output MLT-3 “Dec” signal, based on the comparison. NRZ block  104  (FIG. 4) converts this signal to a suitable binary signal for input to subsequent processing stages, as are known in the art. Alternatively, logic  258  may itself output an NRZ signal, so that an additional NRZ block is not needed. 
     Returning to FIG. 11, the output “Dec” signal is also input to adders  260  and  262 , and as a control signal to a multiplexer  264 . Adders  260  and  262  subtract the incoming equalized signals via junction  266  from the “Dec” signal. Multiplexer  264  selects among the outputs of adders  260  and  262  and the incoming equalized signal, depending upon the “Dec” output of logic  258 , and outputs an “Error” signal dependent on the difference between the input equalized signal and the control “Dec” signal. The Error signal will be driven to zero when the equalized signal input levels stabilize at the appropriate 1, 0 and −1 levels. 
     It will be appreciated that other arrangements of the modules described hereinabove may also be used advantageously in other receivers. All such arrangements, and their use in receiving digital signals, are considered to be within the scope of the present invention. The principles of the present invention thus enable receivers to receive data with superior accuracy and reduced symbol error, compared to receivers at present known in the art. 
     It will be further appreciated that the preferred embodiments described above are cited by way of example, and the full scope of the invention is limited only by the claims.