Abstract:
High-performance, digital-to-analog conversion (DAC) suitable for use in systems implemented with low-voltage, low-power integrated circuit fabrication processes is disclosed. Encoder circuitry receives a binary number for which an analog representation is sought. Segments of the binary number are thermometer encoded and complemented to provide signals to drive analog conversion circuitry. The analog conversion circuitry includes sets of current cells, with each cell in a set contributing an equal amount to one or the other of the complementary legs of the analog output of the converter. Each current cell is a fully differential current switch with charge canceling, fed by a regulated cascode current source. The regulated cascode current source offers uncharacteristically high impedance that contributes to good circuit performance even in low-voltage, low-power implementations. Other design factors of the current cell contribute significantly to overall performance. Hierarchical gradient symmetry cancellation techniques are also employed to reduce integral non-linearity attributable to process-related surface gradients.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation of application Ser. No. 09/837,928, filed Apr. 19, 2001 now U.S. Pat. No. 6,407,688, which is a continuation of application Ser. No. 09/383,068, filed Aug. 25, 1999, now issued as U.S. Pat. No. 6,295,012, the disclosure of which is incorporated fully herein by reference. 
    
    
     BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The invention relates to electronic devices and in particular, to those employing digital-to-analog conversion circuitry. 
     2. Description of Related Art 
     Modern electronic systems are typically realized as a complete “system on a chip.” Such systems typically integrate analog and digital functionality onto the die of a single integrated circuit. Such systems offer lower cost, power, and size benefits to the customer. 
     Often a system will be based on a digital signal processing (DSP) core that implements system functionality through the use of discrete mathematical algorithms that are realized through hardware, firmware, or programmable means. In order for the system to interface to analog based continuous signals, such systems typically employ the use of a digital-to-analog converter (DAC). Examples of such systems include direct digital synthesis (DDS) products, TDMA/CDMA wireless communication systems, as well as audio and video devices. 
     CMOS continues to be the dominant process used to fabricate integrated circuits that contain such systems-on-a-chip. Driven by the desire for further miniaturization, advances in CMOS fabrication processes continue to lead to integrated circuits with lower and lower operating voltage and power specifications. While digital circuit designs can readily be transported to a more advanced process, analog circuit designs often produce poorer results when transported, or cannot be transported at all. 
     Traditional circuit designs for digital-to-analog converters suffer in this respect and generally perform poorly when moved to advanced CMOS fabrication processes. Consequently, there is a need in the art for a digital-to-analog converter providing both good AC and DC performance characteristics, and occupying minimal die space, when implemented using advanced integrated circuit fabrication processes. 
     SUMMARY OF THE INVENTION 
     The invention may be employed to provide high-performance digital-to-analog conversion suitable for use in systems implemented with low-voltage, low-power integrated circuit fabrication processes. The digital-to-analog converter embodiment described herein includes encoder circuitry and analog conversion circuitry. The encoder circuitry receives a binary number for which an analog representation is sought. Segments of the binary number each feed into a binary-to-thermometer encoder. Each binary-to-thermometer encoder turns on the number of output signals that corresponds to the value represented at its inputs. Latch elements latch the output signals of each binary-to-thermometer encoder, and present each signal and its complement as outputs to the analog conversion circuitry. 
     The analog conversion circuitry includes a set of current switching cells for each segment of the binary number fed to a binary-to-thermometer encoder. Each cell in a set contributes an equal amount to the analog output of the converter. Each cell is controlled by one of the output signals of the encoder circuitry latches and its complement, to contribute its total weight to one or the other of the complementary outputs of the converter. 
     Each current cell is a fully differential current switch with charge canceling, fed by a regulated cascode current source. The regulated cascode current source receives its input current from a master curr_ent bia_s_cj_tcuit_through a pair of mirror transistors. The regulated cascode current source offers uncharacteristically high impedance that contributes to good circuit performance even in low-voltage, low-power implementations. Other design factors of the current cell contribute significantly to overall performance. 
     Hierarchical gradient symmetry cancellation techniques are employed to assign switching order assignments to the cells within each set in order to reduce integral non-linearity attributable to process-related surface gradients. 
     These and other purposes and advantages of the present invention will become more apparent to those skilled in the art from the following detailed description in conjunction with the appended drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram of a digital-to-analog converter. 
     FIG. 2 is a schematic diagram of a master current bias circuit. 
     FIG. 3 is a schematic diagram of a current cell circuit. 
     FIG. 4 illustrates the layout orientation of current cell matrices and the relative switching order among the cells in each matrix. 
    
    
     In the figures just described, like parts appearing in multiple figures are numbered the same in each figure. 
     DETAILED DESCRIPTION 
     The present invention provides improved circuit design for electronic devices requiring low-voltage digital-to-analog conversion. In the following description, numerous details are set forth in order to enable a thorough understanding of the present invention. Many such details relate to an embodiment of the present invention using a 3.3 Volt CMOS fabrication process to implement digital-to-analog conversion for a 10-bit binary value. However, it will be understood by those of ordinary skill in the art that these specific details are not required in order to practice the invention. Further, well-known elements, devices, process steps and the like are not set forth in detail in order to avoid obscuring the present invention. 
     FIG. 1 is a block diagram of a digital-to-analog converter (DAC). Digital-to-analog converter  100  comprises encoder circuitry  110 , and analog conversion circuitry  160 . Encoder circuitry  110  further comprises data register  120 , binary-to-thermometer encoders  130 ,  132 , latches  140 ,  142 , inverter  112 , and interconnecting signal pathways  122 ,  124 ,  134 ,  136 ,  150 ,  152 ,  154 ,  156 . Analog conversion circuitry  160  further comprises external bias current connection  161 , master current bias circuitry  162 , current cell matrices  170 ,  172 , signal pathways  180 ,  182 ,  184 ,  186 , and current summing nodes  190 ,  192 . 
     Encoder Circuitry 
     Encoder circuitry  110  functions to receive a binary number at its input  101  and to present complementary signal pairs at its outputs  150 - 156  that are representative of the received binary number. The binary number is the digital value for which the DAC is to produce an analog counterpart. The complementary pairs will be used by the analog conversion circuitry to turn on and off individual cells that contribute to the complementary analog output of the DAC. 
     Operation of the various circuit elements within encoder circuitry  110  are synchronized by means of a clock signal presented at input  102 . Data register  120  temporarily stores the incoming binary number at the rising edge of the clock signal. In the presently described embodiment, data register  120  is a 10-bit register, permitting the binary number to have a maximum value of 21°−1, or 1023 (i.e., 1024 possible values including zero). 
     Upon storage by the data register  120 , the 10 bits representing the binary number are communicated over signal pathways  122 ,  124  to binary-to-thermometer encoders  130 , 132 . Binary-to-thermometer encoders  130 ,  132  are also referred to as unit encoders, herein. Each unit encoder operates such that the number of bits turned on at its output is the same as the numeric value presented at its input. The numeric value presented at its multi-bit input is presumed to be a binary number. A binary number has a least significant bit with a unit value (i.e., 20=1), and each successive bit represents twice the numeric value of the preceding bit. When the number is unit coded all bits share the same significance, i.e., the unit value. 
     A least significant segment of the binary number stored in register  120 , comprising the four least significant bits (LSB), are communicated to LSB unit encoder  132  over signal pathway  124 . Signal pathway  124  communicates four bits in parallel. LSB unit encoder  132  converts the 4-bit input to a 16-bit (24=16), unit coded output. Note that one of the 16 output bits will always be in the off state and is included for design convenience. While the 4-bit input can represent 16 possible values, one of those values is zero. Accordingly, the highest numeric value that can be represented is 15, so at most 15 unit coded outputs will be in the on state. 
     A most significant segment of the binary number stored in register  120 , comprising the six most significant bits (MSB), are communicated to MSB unit encoder  130  over signal pathway  122 . Signal pathway  122  communicates six bits in parallel. MSB unit encoder  130  converts the 6-bit input to a 64-bit (26=64), unit coded output. Note that one of the 64 output bits will always be in the off state and is included for design convenience. While the 6-bit input can represent 64 possible 
     values, one of those values is zero. Accordingly, the highest numeric value that can be represented is 63, so at most 63 unit coded outputs will be in the on state. 
     Each of unit encoders  130 ,  132  performs and completes the encoding operation during the “on” state of the master clock signal that first triggered the storage of a binary number in register  120 . Completion of the encoding operation includes presenting outputs in a settled and static state. In the described embodiment, unit encoders  130 ,  132  were coded in the Verilog programming language and synthesized into transistor-based hardware. 
     Unit encoder  130  communicates its output to MSB latch circuitry  140  over signal pathway  134 . Signal pathway  134  communicates 64 bits in parallel. Unit encoder  132  communicates its output to LSB latch circuitry  142  over signal pathway  136 . Signal pathway  136  communicates 16 bits in parallel. 
     MSB latch circuitry  140  and LSB latch circuitry  142  function as “slave” registers. The clock signal that drives these registers is an inverted version of the master clock signal that drives register  120 . The inversion of the master clock signal is implemented through simple inverter  112 . MSB latch circuitry  140  and LSB latch circuitry  142  store their input signals communicated over signal pathways  134  and  136 , respectively, on the rising edge of the inverted master clock signal. The MSB  140  and LSB  142  latch circuitry present stable output values to analog conversion circuitry  160  for a first binary number during the time a second binary number is being stored by register  120  and encoded by unit encoders  130 ,  132 . 
     The MSB  140  and LSB  142  latch circuitry further produce a complementary pair of outputs for each bit of their inputs. The first output signal of the complementary pair is identical in value to the corresponding input bit. The second output signal of the complementary pair is the complement, or inverse, of the value of the corresponding input bit. MSB latch circuitry  140  communicates the non-inverted signals of the complementary pairs on signal pathway  150 , and the inverted signals of the complementary pairs on signal pathway  152 , to the MSB current cell matrix  170 . LSB latch circuitry  142  communicates the non-inverted signals of the complementary pairs on signal pathway  154 , and the inverted signals of the complementary pairs on signal pathway  156 , to the LSB current cell matrix  172 . 
     One complementary output pair of MSB latch circuitry  140  corresponds to the single bit in the output of MSB unit encoder  130  that is always in the off state as described earlier. Similarly, one complementary output pair of LSB latch circuitry  142  corresponds to the single bit in the output of LSB unit encoder  132  that is always in the off state. In some embodiments such an output pair corresponding to an always-off input is not communicated to the analog conversion circuitry. Other embodiments may communicate such an output pair to the analog conversion circuitry in modified form; i.e., the outputs are not complementary but are both fixed in the off state. One skilled in the art recognizes that these and other embodiments may be employed in the practice of the invention. 
     Analog Conversion Circuitry 
     The analog conversion circuitry  160  prominently contains a plurality of current cells. Differential outputs from each of the current cells are summed at current summing nodes  190 ,  192 , and drive a pair of load resistors that are centered at ground (not shown). The current driving the load resistors is related to the full-scale current in almost identical proportion as the original binary number relates to the maximum binary number the DAC  100  accommodates, i.e., 1023. Thus, the current represents an analog approximation of the original number in binary (digital) form. 
     The individual current cells obtain current from master current bias circuit  162 . FIG. 2 is a schematic diagram of the master current bias circuit used in the present embodiment. The master current bias circuit  162  uses a pair of low voltage, wide swing, high impedance current mirrors. The mirrors receive current from some stable current source at the external current bias connection point  161 . The stable current source may be, for example, a bandgap reference current tap. The mirrors take in the current source and translate it to a current sink for use by the individual current cells. The current mirror and the bandgap reference current tap are well known in the art. 
     The presently described embodiment organizes the plurality of current cells into two sets of current cells, each physically configured as a matrix. MSB current cell matrix  170  is controlled by the complementary output signal pairs of MSB latch circuitry  140 . LSB current cell matrix  172  is controlled by the complementary output signal pairs of LSB latch circuitry  142 . 
     The MSB current cell matrix  170  is a coarse conversion matrix. In the presently described embodiment the MSB matrix  170  is fabricated on an integrated circuit die in an eight column by eight row matrix configuration, providing 64 cells. One sixty-fourth ({fraction (1/64)}={fraction (1/26)}) of the nominal full-scale DAC output current is distributed equally to each of 63 of the 64 current cells in MSB matrix  170 . One of the 64 current cells remains unused. The unused one sixty-fourth of the full-scale current supplies all of the current cells in the LSB current cell matrix  172 . The full-scale current of the present embodiment is on the order of 20 milliamps. 
     Each of the cells is controlled by one of the complementary output signal pairs of MSB latch circuitry  140  to deliver its portion of the full-scale current to one or the other of the current summing nodes  190 ,  192  at the output of the DAC  100 . An exception in the present embodiment is the unused current cell. Its control input signals are not complementary, but rather are both fixed in the off state. This prevents the cell from making a contribution to either summing node. 
     The LSB current cell matrix  172  is a fine conversion matrix. In the presently described embodiment the LSB matrix  172  is fabricated on an integrated circuit die in an eight row by two column matrix configuration, providing 16 cells. One sixteenth ({fraction (1/16)}={fraction (1/24)}) of the one sixty-fourth of the nominal full-scale current unused by the MSB current cell matrix  170  is distributed equally to each of 15 of the 16 current cells in LSB matrix  172 . One of the 16 current cells remains unused to accommodate a zero value. (The unused {fraction (1/1024)} ({fraction (1/24)}+6) of the nominal full-scale current remains unused resulting in an operational full-scale current just slightly below the nominal value.) 
     In similar fashion to the MSB cell matrix  170 , each of the cells in the LSB current cell matrix  172  is controlled by one of the complementary output signal pairs of LSB latch circuitry  142  to deliver its portion of the full-scale current to one or the other of the current summing nodes  190 ,  192  at the output of the DAC  100 . As above, an exception is the unused current cell. Its control input signals are not complementary, but rather are both fixed in the off state. This prevents the cell from making a contribution to either summing node. 
     Employing such equal current-based weighting to each current cell in a matrix provides faster settling time than with, for example, voltage divider DAC designs. This improves AC and DC performance and represents an advantage of the present invention. 
     Current Cell Detail 
     FIG. 3 is a schematic diagram of a current cell circuit. The MSB matrix ( 170  of FIG. 1) and the LSB matrix ( 172  of FIG. 1) both employ the current cell architecture represented in the schematic for their individual current cells. Because a current cell in the MSB matrix  170  conducts some multiple of the current conducted by an LSB matrix  172  current cell, however, transistor sizes are scaled accordingly. 
     Each current cell  300  as depicted in FIG. 3 is a fully differential current switch, which takes in a pair of signals having complementary binary states, and passes a differential signal out to two loads having a common DC reference. 
     Each current cell  300  comprises a differential current switch circuit  310 , a regulated cascode current source  320 , a master current bias slave circuit  325 , a DC reference voltage connection  370 , inputs for a complementary signal pair  350 ,  352 , differential outputs  354 ,  356 , electrical ground connection  378 , and master current bias connections  372 ,  374 ,  376 . The differential current switch  310  further comprises current source connection  312 , switching transistors  330 ,  332 , and charge canceling transistors  334 ,  336 . The regulated cascode current source  320  further comprises cascode transistors  340 ,  342 , DC bias node  341 , impedance multiplier transistor  344 , and capacitance element  346 . The master current bias slave circuit  325  further comprises master current supply bias mirror transistors  360 ,  362 . 
     Differential current switch  310  receives current from the regulated cascode current source  320  at current source connection  312 . The source of each of PMOS switching transistors  330 ,  332  is connected to current source connection  312 . The drain of switching transistor  330  is connected to the source of charge canceling transistor  334 . The source and drain of charge canceling transistor  334  are short-circuited, and the drain is further connected to one of the differential outputs  356 . Switching transistor  330  has its gate connected to an input  350  for one of the signals of a complementary pair. Charge canceling transistor  334 , in contrast, has its gate connected to an input  352  for the complementary signal of the pair. 
     In similar, but complementary, fashion, the drain of switching transistor  332  is connected to the source of charge canceling transistor  336 . The source and drain of charge canceling transistor  336  are short-circuited, and the drain is further connected to the remaining differential output  354 . Switching transistor  332  has its gate connected to input  352  for one of the signals of the complementary pair. Charge canceling transistor  336  has its gate connected to an input  350  for the complementary signal of the pair. 
     Accordingly, it can be seen that the gates of the switching transistors  330 ,  332  are driven by complementary signals, as are the gates of the charge canceling transistors  334 ,  336 . 
     Each of charge canceling transistors  334 ,  336  is roughly equal to one half the size of its corresponding switching transistor, i.e.,  330 ,  332 , respectively. The charge canceling transistors  334 ,  336  cancel unwanted channel charge injection and minimize unwanted clock feed-through from the gate stimulus, by canceling charges between the switch transistor and the complementarily switched charge canceling transistor. This configuration achieves minimal unwanted feed-through to the loads coupled to the differential outputs  354 ,  356 . The reduced feed-through minimizes harmonic distortion, improving spurious free dynamic range (SFDR). This represents a further advantage of the present invention. 
     Regulated cascode current source  320  delivers current to the differential current switch  310  at current source connection  312 . Regulated cascode current source  320  uses a very high impedance cascode configuration (e.g., 100 Megaohms) to source up to the full value of the current through either leg of the differential current switch  310  depending on the value of the complementary input code. The cascode configuration employs two series cascoded transistors  340 ,  342 . The source of cascode transistor  340  is connected to a common DC reference voltage connection  370 . The drain of cascode transistor  340  is connected to the source of cascode transistor  342  at DC bias node  341 . The drain of cascode transistor  342  is connected to current source connection  312 . 
     By utilizing a regulated cascode current source configuration the stacked transistor area can be kept smaller than with a conventional stacked cascode configuration, while still maintaining high output impedance. The smaller area also serves to impair either active switching signal from feeding through the drain-source path of the cascode transistor  342  and ultimately to DC bias node  341 . This improves differential non-linearity (DNL) characteristics by reducing disturbance at DC bias node  341 . This represents a further advantage of the present invention. 
     The DC bias node  341  at which both cascode transistors  340 ,  342  meet is connected to the gate of impedance multiplier transistor  344 . The drain of transistor  344  is fed back to the gate of cascode transistor  342 . The source of transistor  344  is connected to a common DC reference voltage connection  370 . 
     Impedance multiplier transistor  344  operates to effectively multiply the high impedance of the cascode output by a factor of the gain of the transistor  344 . Transistor  344  further adds an additional path for unwanted charge at connection  312  to travel, and so reduces unwanted signal injection to the common DC bias node  341 . Capacitance element  346  is connected in parallel with impedance multiplier transistor  344 , increasing transient stability while further minimizing unwanted signal feed-through to the DC bias node  341 . So increasing the stability of common DC bias node  341  reduces glitch energy transferred to the loads coupled to the differential outputs  354 ,  356 , again reducing harmonic distortion, and improving SFDR. These operational characteristics represent yet another advantage of the present invention. 
     Further, the higher impedance of the individual cells than seen in earlier DAC designs helps maintain a higher overall impedance as seen by the load when the differential outputs of all the current cells are connected in parallel to the current summing nodes. The higher impedance contributes to lower integral non-linearity (INL) characteristics, a further advantage of the present invention. 
     FIG. 4 illustrates the layout orientation of current cell matrices and the relative switching order among the cells in each matrix. As described earlier, the MSB current cell circuitry  170  is configured as a matrix of 8 rows by 8 columns. The number appearing within each cell of the MSB matrix  170  in FIG. 3 indicates the switching order of the cell. In accordance with the unit encoding described earlier in relation to the encoder circuitry ( 110  of FIG.  1 ), for any cell switched on within the MSB matrix  170 , all other cells within the MSB matrix  170  having a switching order number lower than that of the switched on cell, will also be switched on. The complementary signal pairs communicated from encoder circuitry ( 110  of FIG. 1) via signal paths ( 150 ,  152  of FIG. 1) are connected to the individual current cells of MSB matrix  170  to produce the switching order depicted in FIG.  4 . Cell  410  is the unused cell of the matrix  170  as described above in reference to FIG.  1 . 
     The LSB current cell circuitry  172  is configured as a matrix of 8 rows by 2 columns. The number appearing within each cell of the LSB matrix  172  in FIG. 3, similarly, indicates the switching order of the cell. As with the MSB matrix  170 , for any cell switched on within the LSB matrix  172 , all other cells within the LSB matrix  172  having a switching order number lower than that of the switched on cell, will also be switched on. The complementary signal pairs communicated from encoder circuitry ( 110  of FIG. 1) via signal paths ( 154 ,  156  of FIG. 1) are connected to the individual current cells of LSB matrix  172  to produce the switching order depicted in FIG.  4 . Cell  412  is the unused cell of the matrix  172  as described above in reference to FIG.  1 . 
     The cell switching order assignments depicted in FIG. 4 utilize hierarchical gradient symmetry cancellation techniques. Such a layout orientation has the advantage of reducing INL attributable to process-related surface gradients. This represents a further advantage of the present invention. 
     Various modifications to the preferred embodiment can be made without departing from the spirit and scope of the invention. For example, the design could be extended or contracted to accommodate a binary input number having more or fewer than 10 bits. Thus, the foregoing description is not intended to limit the invention which is described in the appended claims in which: