Abstract:
Improved methods and systems for feedback signals in a sensor system. An example method demodulates a sense signal using an analog demodulator and also demodulates the sense signal using a digital demodulator. The difference between the result of the analog demodulator and the digital demodulator is determined and then integrated. A sensor feedback control signal is generated based on the integrated difference.

Description:
BACKGROUND OF THE INVENTION 
   Closed loop sensing systems, such as accelerometers and gyroscopes, that include synchronous detection perform the best when they include a feedback loop with relatively high bandwidth, high dynamic range and high precision. Typical analog feedback loops provide high bandwidth and dynamic range, but have limited precision due to analog errors, such as offset voltages and demodulator imperfections. Common high bandwidth digital feedback loops with digital demodulators and accumulators offer high precision but exhibit lower dynamic range. High speed digital to analog converters (DACS) are typically limited to about  16  bits of resolution. DACs with higher resolution exists, but at the expense of lower bandwidth. 
   Therefore, there exists a need for high bandwidth, high dynamic range and high precision feedback loops used in sensing systems. 
   SUMMARY OF THE INVENTION 
   The present invention provides improved methods and systems for feedback signals in a sensor system. An example method demodulates a sense signal using an analog demodulator and also demodulates the sense signal using a digital demodulator. The difference between the result of the analog demodulator and the digital demodulator is determined and then integrated. A sensor feedback control signal is generated based on the integrated difference. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     Preferred and alternative embodiments of the present invention are described in detail below with reference to the following drawings: 
       FIGS. 1-4  illustrate examples of systems formed in accordance with various embodiments of the present invention. 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     FIG. 1  shows an example system  20  that includes a sensing element  24 , analog front-end circuit components  36 , an analog-to-digital (A/D) converter  38 , an analog demodulation component  30 , a digital signal processing component  32 , a digital-to-analog converter (DAC)  40 , an operational amplifier (Op Amp)  50 , an analog integrator  52 , a sensing element drive  54 , a modulation generator  46 , and a comparator  44 . The analog demodulation component  30  includes an analog demodulator  60  and a low pass filter  62 . The digital signal processing component  32  includes a demodulator  64 , an accumulator  68 , and a phase shifter  66 . The resonator fiber optic gyro (RFOG) ( 24 ) is a rotation sensor that may employ this type of signal processing and control. For the RFOG, the sensing element drive  54  controls a laser frequency to a resonance frequency of a fiber ring resonator. The modulation generator  46  modulates the laser frequency, which is converted to a light intensity signal by the resonator. The light intensity signal is converted to a voltage signal by a photodetector (not shown). The voltage detector signal passes through the analog front-end  36  to be demodulated by analog demodulator  60  and digital demodulator  64 . The output of the analog demodulator  60  is a combination of a resonance error signal and demodulator errors. The resonance error signal is approximately proportional to the difference between the laser frequency and the fiber ring resonator resonance frequency for a frequency range of about half the resonator resonance width away from the resonance frequency. For an ideal resonator the resonance error signal is zero when the laser frequency is at the resonator resonance frequency. To control the laser to the resonator resonance frequency, the analog integrator  52  integrates the error signal and adjusts the laser frequency until the error signal is driven to zero value. However, imperfections in the analog demodulator  60  and analog integrator  52  result in a dc offset error at the integrator input. The dc offset is integrated by the integrator  52 , which adjusts the laser frequency slightly off the resonance frequency until the resonance error signal has an amplitude and sign such that it cancels out the dc offset error. The laser frequency deviation from the resonance frequency can result in rotation sensing errors. The digital demodulator  64  also outputs a digital value that is approximately proportional to the frequency difference between the laser frequency and the resonator resonance frequency. Unlike the analog demodulator  60  and analog integrator  52 , the digital demodulator  64  and digital accumulator  68  do not generate a dc offset error. The digital accumulator  68  and DAC  40  adjusts an analog signal at the Op Amp  50  until the digital demodulator  64  output is driven to zero, which only occurs when the laser frequency is at the resonance frequency. Therefore the digital feedback loop provides an analog correction signal that cancels out the dc offset error generated by the analog demodulator and analog integrator. Since changes in the dc offset error are caused mostly by aging and temperature effects, the change in dc offset error with time is very slow, typically much less than 1 Hz. Therefore, the digital correction loop does not need to be a high bandwidth signal to correct for analog errors. Since the bandwidth requirement on the digital loop is very low (below 1 Hz) the DAC  40  can be of the type that provides very high resolution, such as a σ-Δ DAC. High resolution of DAC  40  may be necessary to cancel out analog errors to a required level of accuracy. 
   The sensing element  24  outputs a sensing signal to the analog front-end circuit components  36 , which process the sensing signal and sends it to the A/D converter  38  and the analog demodulator  60  in the analog demodulation component  30 . The analog demodulator  60  demodulates the signal based on a signal from a local oscillator which is a phase shifted signal that is the output of the phase shifter  66  in the digital demodulation component  32 . The phase shifter  66  receives the output of the comparator  44 . The comparator  44  compares the modulation signal generated by the modulation generator  46  to some threshold voltage, for example zero volts. The output of the comparator  44  is a digital signal which transitions to a high or low state when the modulation signal passes through a threshold level. The modulation signal is also sent to the sensing element  24 . The output of the analog demodulator  60  is filtered by the low pass filter  62 . 
   The demodulator  64  demodulates the digital signal generated by the A/D converter  38  based on the signal from a local oscillator which is a phase shifted signal that is the output of the phase shifter  66 . The accumulator  68  receives the output of the demodulator  64 . The DAC  40  receives the output of the accumulator  68 . 
   The Op Amp  50  receives the output of the low pass filter  62  and subtracts the DAC  40  output signal from the low pass filter  62  output signal. The analog integrator  52  integrates the difference determined by the Op Amp  50  to produce an adjusted error signal. The sensing element drive  54  generates a drive signal for the sensing element  24  based on the adjusted error signal from the Op Amp  50 . The Op Amp  50  has an input offset voltage that will cause the analog integrator  52  to swing towards saturation, even if the error signal is at zero. However, as long as the offset voltage is small enough, the integrator output will change the laser frequency to be slightly off resonance in a way to generate an error signal that exactly cancels out the Op Amp offset voltage and thus maintains the integrator  52  in stable operation. Because of an Op Amp input offset voltage, the laser frequency will not be exactly on the resonance frequency, which will lead to a rotation sensing error. There are other error sources in the analog front-end and analog mixer that can have the same effect, and can be represented as an effective integrator input offset error. The digital demodulator  64  has much smaller errors than the analog demodulator  60  (both perform the same function of demodulation) and the digital accumulator  68  has much smaller errors than the analog integrator  52  (both perform the same function of integration). The accumulator  68  adjusts the DAC output until the digital demodulator  64  output is zero, which occurs when the laser frequency is at the resonator resonance frequency. Therefore, the DAC output is adjusted to exactly cancel out the analog errors such as integrator input offset. 
     FIG. 2A  shows a system  100  similar to system  20 . The system  100  includes a digital demodulation component  104  that includes all the components of the digital demodulation component  32  but with a random noise generator  108  and an adder  110 . The adder  110  adds a random number signal generated by the random noise generator  108  to the output of the digital accumulator  68 . The DAC  40  receives the output of the adder  110 . A low pass filter  114  filters the analog signal generated by the DAC  40 . The random number signal is noise that improves the resolution of the DAC  40 . The accumulator  68  can provide an output value with the desired resolution. However, standard low cost DACs may not provide the required resolution. By adding noise to the accumulator output value the effective resolution of the DAC can be significantly improved. The amplitude of the noise must be roughly a few bits of the DAC so that the DAC output is dithered about the accumulator output value. The output of the DAC for each DAC sample is still quantized to the DAC least significant bit (LSB). However, if many DAC samples are averaged together, the quantization of the average value is smaller than the DAC LSB. The level of quantization of the average value is reduced by increasing the number of DAC samples in the average. By increasing the DAC sample frequency the quantization of the average value can be decreased in a fixed time period. Therefore this technique requires a DAC that can operate at a much higher frequency than the bandwidth of the digital control loop in order to have enough DAC samples in an averaging time that does not impact the digital loop control bandwidth. The low pass filter  114  essentially averages the DAC output. If the random noise has frequency components that are within the digital loop bandwidth, then the random noise will contribute undesirable noise into the sensor. The random noise generator  108  can be made to provide noise that has frequency components only above the loop bandwidth and above the cut-off frequency of the low pass filter  114 , which removes the noise before being added to the analog control loop. 
     FIG. 2B  illustrates a system  100 a that includes the features of the system  100  but is applied to a Resonator Fiber Optic Gyro Laser Drive architecture. The system  100   a  includes an integrator  52   a  and another integrator  52   b  with a lead-lag circuit for employing a second order feedback loop, which is commonly used to provide a stable loop with very high bandwidth. A summing amplifier  120  receives the output of the second integrator  52   b  and outputs a signal to a laser drive component  54   a.  The summing amplifier  120  also receives the output of the modulation generator  46 , which is a Direct Digital Synthesizer (DDS). The laser drive  54   a  provides both the modulation signal to detect the center of the resonator resonance feature and the control signal to lock the laser onto the resonance frequency. 
     FIG. 3A  illustrates another system  130  that includes a digital demodulation component  132  that includes all the components of the digital demodulation component  32  but with a σ-Δ modulator  134 . The σ-Δ modulator  134  receives the output of the digital accumulator  68 . The σ-Δ modulator  134  combined with DAC  40  increases the resolution of DAC  40  beyond the bit resolution of the DAC, allowing lower cost and lower power DACs to be used. The σ-Δ modulator does generate noise, but with a noise spectrum that decreases with decreasing frequency. The low pass filter  114  removes most of the noise generated by the σ-Δ modulator  134  before reaching the laser drive. 
     FIG. 3B  shows a system  130   a  that is the system  130  that includes an integrator  52   a  and another integrator  52   b  with a lead-lag circuit for employing a second order feedback loop, with the digital demodulation component  132  and is used in a Resonator Fiber Optic Gyro Laser Drive architecture. 
     FIG. 4A  illustrates another system  160  that includes a digital demodulation component  162  that includes all the components of the digital demodulation component  32  but with a pulse width modulator (PWM)  164 . The PWM  164  receives the output of the digital accumulator  68 . The pulse width of the output of PWM is proportional to the accumulator  68  output value. The low pass filter  114  averages the output signal from the PWM  164 , thus converting the digital output to an analog signal. Because the PWM  164  and low pass filter  114  produces an analog signal, a DAC is not required, thus eliminating an active component and simplifying the design. 
     FIG. 4B  shows a system  160   a  that is the system  160  that includes an integrator  52   a  and another integrator  52   b  with a lead-lag circuit for employing a second order feedback loop, with the digital demodulation component  162  and is used in a Resonator Fiber Optic Gyro Laser Drive architecture. 
   Low frequency wandering on the output signal of the analog loop (the analog demodulation component  30 ) is due to analog drift and temperature errors that typically occur at frequencies much less than 1 Hz. The higher frequency variation on the output signal of the analog loop is due to noise in the electronics, laser and resonator and finite bandwidth of the fast analog loop. If the digital loop is too slow, then it will not be able to follow the wandering part of output signal of the analog demodulation component  30 , and the output signal of the DAC  40  or the digital loop will have some phase lag and lower amplitude, thus will not perfectly cancel out analog errors. Therefore, the digital loop (digital demodulation component  32 ) must be fast enough so that it can follow the drift and temperature induced analog errors to an acceptable level. If the digital loop is fast enough, the output signal of the DAC  40  or digital loop should look just like the lower frequency wandering part of the output signal of the analog loop and should be in phase. In one embodiment, the bandwidth of the fast analog loop is between 1 kHz and 1 MHz and the bandwidth of the digital loop is a few hertz or less to adequately follow the drift and temperature induced errors. 
   While the preferred embodiment of the invention has been illustrated and described, as noted above, many changes can be made without departing from the spirit and scope of the invention. Accordingly, the scope of the invention is not limited by the disclosure of the preferred embodiment. Instead, the invention should be determined entirely by reference to the claims that follow.