Abstract:
A linearization circuit includes a sensor circuit having a first terminal receiving an excitation voltage, and second and third terminals producing a sensor output voltage therebetween. A differential amplifier circuit produces a linearization current, and a scaling circuit operates to produce a scaled linearization current in response to the linearization current. A current direction switch circuit includes a fourth terminal receiving the scaled linearization current, a fifth terminal and conducting a correction current proportional to the linearization current, and a control terminal receiving a polarity control signal to determine the direction of flow of the correction current through the fifth terminal in response to the sensor output voltage. An amplifier circuit receives and amplifies a reference voltage to produce the excitation voltage, the amplifier circuit including a feedback circuit, the feedback circuit being coupled to the fifth terminal and modulating the feedback circuit in response to the correction current to cause the amplifier circuit to produce the excitation voltage equal to the reference voltage plus or minus a positive or negative correction, respectively, according to the level of the polarity control signal and according to the magnitude of the sensor output voltage.

Description:
BACKGROUND OF THE INVENTION 
     The invention relates to a circuit for precisely correcting positive and negative linearity errors of a voltage-excited bridge sensor by a technique utilizing a minimum amount of circuitry and a minimum number of external package leads. 
     Resistive bridge circuits, i.e., bridge sensors, have nonlinearities due to mismatches in values of the bridge circuit elements. Many bridge sensors are inherently non-linear. It is possible to compensate for such non-linearity by varying the bridge excitation voltage proportionally to the output unbalance signal of the bridge. The following equation represents the bridge excitation voltage V EXCITE : 
     
       
           V   EXCITE   =V   EXCITE(0)   ±V   BROUT   ×K   LIN ,  (Equation 1) 
       
     
     where V BROUT  is the bridge circuit output voltage, K LIN  is a linearization constant, and V EXCITE(0)  is an initial value of V EXCITE . 
     The uncorrected signal results in a non-linear curve for V BROUT , as indicated by curve A in FIG.  5 . Curve B in FIG. 5 represents the usually parabolic relative non-linearity of the bridge transducer that results in the nonlinear output of the bridge circuit indicated by curve A. Curve C represents the non-linearity after correction or linearization by varying the excitation voltage V EXCITE , and curve D represents the corrected bridge output voltage obtained as a result of correcting the excitation voltage by means of a feedback circuit coupled between the bridge output and V EXCITE . 
     A very effective technique for “linearizing” a bridge circuit is to modulate its “excitation source”, i.e., the reference voltage which is applied to the bridge circuit. U.S. Pat. Nos. 4,190,796, 4,362,060, 4,492,122, 5,122,756 and 5,764,067 are illustrative of the state of the art. The known linearization circuits generally are used in conjunction with conventional instrumentation amplifiers which provide amplified outputs to suitable utilization circuits. 
     The above mentioned known linearization circuits generally require four external package leads to allow a user to determine both the polarity and magnitude of linearity corrections required for each individual bridge sensor circuit. However, the user often has no way of knowing in advance whether the polarity of linearity correction needed for a particular bridge sensor circuit is positive or negative. Consequently, the user may have to swap connections between two external leads of the bridge linearization circuit to get the correct polarity of linearization correction, which is inconvenient. Furthermore, it usually is undesirable to have to use more external package leads than is genuinely necessary, and it would be better to be able to adjust the magnitude of the needed correction with one, rather than two external package leads. 
     Accordingly, there is an unmet need for an improved bridge linearity correction technique which requires a reduced amount of circuitry and a reduced number of external package leads for setting both the polarity and magnitude of the linearity corrections required for each different bridge sensor. 
     SUMMARY OF THE INVENTION 
     Accordingly, it is an object of the invention to provide a bridge sensor linearization circuit and technique for providing a correction to the excitation voltage of a bridge sensor circuit using a reduced number of circuit components. 
     It is another object of the invention to provide a bridge linearization circuit and technique for providing a correction in the bridge excitation voltage of the desired polarity and magnitude requiring only two external package leads, one to select the polarity of the needed correction and the other to establish the magnitude of the needed correction. 
     It is another object of the invention to avoid the need to swap package lead connections to establish the correct polarity of a correction to a bridge excitation voltage produced by a linearization circuit. 
     It is another object of the invention to avoid the need for a user to construct “build-your-own” circuitry to obtain the needed linearization of a bridge sensor circuit. 
     It is another object of the invention to avoid dependance of the linearization constant K LIN  on variations of absolute resistances of on-chip integrated circuit resistors. 
     Briefly described, and in accordance with one embodiment thereof, the invention provides a linearization circuit including a sensor circuit having a first terminal receiving an excitation voltage, and second and third terminals producing a sensor output voltage therebetween, a differential amplifier circuit coupled to the second and third terminals and producing a linearization current in response to the sensor output voltage, a current direction switch circuit producing a bi-directional correction current proportional to the linearization current, the current direction switch circuit having a fourth terminal receiving the linearization current, a fifth terminal conducting the correction current, and a control terminal receiving a polarity control signal to determine the direction of flow of the correction current through the fifth terminal in response to the sensor output voltage, and an amplifier circuit receiving and amplifying a reference voltage to produce the excitation voltage. The amplifier circuit includes a feedback circuit, the feedback circuit being coupled to the fifth terminal and modulating the feedback circuit in response to the correction current to cause the amplifier circuit to produce the excitation voltage equal to the reference voltage plus or minus a positive or negative correction, respectively, according to the level of the polarity control signal and according to the magnitude of the sensor output voltage. 
     In one embodiment, the linearization circuit includes a scaling circuit operative to produce a scaled linearization current in response to the linearization current. The linearization circuit includes a first resistor coupled to the fifth terminal to develop a voltage change on the fifth terminal proportional to the correction current, and further includes a band gap circuit producing the reference voltage. The amplifier circuit includes a differential amplifier having an output coupled to the first terminal, and a feedback resistor coupled between an inverting input of the differential amplifier and the output of the differential amplifier, the inverting input being coupled to the fifth terminal, a non-inverting input of the differential amplifier being coupled to receive the reference voltage. In one embodiment, the current direction switch circuit includes a first switch operatively connecting the fourth terminal to the fifth terminal during a first level of the polarity control signal to conduct the scaled linearization current as the correction current in a first direction through the fifth terminal. A current mirror, a second switch operatively conducts the scaled linearization current through a current mirror control transistor of the current mirror during a second level of the polarity control signal, and a current mirror output transistor of the current mirror producing a replica of the scaled linearization current as the correction current flowing in a second direction through the fifth terminal. 
     In one embodiment of the invention, the differential amplifier circuit includes a first operational amplifier having a non-inverting input coupled to the second terminal, an output coupled to a control terminal of a first output transistor having a first main terminal coupled to a first output conductor and a second main terminal coupled to an inverting input of the first operational amplifier. The inverting input of the first operational amplifier is coupled to a first terminal of a transconductance control resistor. A second operational amplifier includes an inverting input coupled to a second terminal of the transconductance control resistor and to a first main terminal of a second output transistor having a control terminal coupled to an output of the second operational amplifier. The second operational amplifier has a non-inverting input coupled to the third terminal. 
     In one embodiment, the scaling circuit includes an external first resistor coupled between a first external package lead and an external supply voltage. A first differential amplifier includes a non-inverting input coupled to the first external package lead, an inverting input coupled to a first terminal of a second resistor and a first terminal of a transistor having a control terminal coupled to the output of the first differential amplifier. A terminal of the transistor supplies the scaled linearization current through the fourth terminal into the current direction switch circuit, a second terminal of the second resistor being coupled to the external supply voltage. 
     In one embodiment, the scaling circuit includes an MDAC operative to generate the scaled linearization current in response to the linearization current with a scale factor determined by a programmable controller circuit. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a detailed block diagram of a bridge linearization circuit connected to a bridge sensor circuit. 
     FIG. 1A is a schematic diagram of the band gap reference circuit  36  and its connection to amplifier  34  in FIG.  1 . 
     FIG. 2 is a schematic circuit diagram of a current direction switch circuit used in the bridge linearization circuit of FIG.  1 . 
     FIG. 3 is a schematic diagram of an instrumentation voltage-to-current converter in the bridge linearization circuit of FIG.  1 . 
     FIG. 4 is a schematic diagram of a CMOS current direction switch which can be used in the bridge linearization circuit of FIG.  1 . 
     FIG. 5 is a graph useful in explaining operation of the bridge linearization circuit of FIG.  1 . 
     FIG. 6 is a block diagram of an alternative implementation of the system of FIG. 1 utilizing a microprocessor to control scaling of and polarity of the linearization current. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Referring to FIG. 1, bridge linearization circuit  10  includes a bridge sensor circuit  12 , which typically converts an analog quantity such as pressure, strain displacement, light intensity or the like into a low level analog voltage. Bridge sensor circuit  12  typically includes a left arm including resistive elements  12 A and  12 B and a right arm including resistor elements  12 C and  12 D. One or both of resistive elements  12 A and  12 B may be a transducer element. Resistive elements  12 A and  12 C are connected to conductor  14  on which a “bridge excitation voltage” V EXCITE  is produced and modulated in accordance with the present invention. The junction between resistive elements  12 A and  12 B is connected to conductor  18 , and the junction between resistive elements  12 C and  12 D is connected to conductor  16 . The junction between resistive elements  12 B and  12 D is connected to ground. The excitation voltage V EXCITE  is applied to bridge sensor circuit  12  through conductor  14 . The two outputs  16  and  18  of bridge sensor  12  are at identical voltages when bridge circuit  12  is balanced. 
     Alternatively, bridge sensor circuit  12  can be a piezo-resistive semiconductor element which is biased similarly to bridge sensor circuit  12  and produces a similar differential output signal. Therefore, the linearization circuit of the present invention can also correct the non-linearity of such a piezo-resistive semiconductor element. 
     Bridge output conductors  16  and  18  are connected to the (+) and (−) inputs, respectively, of an instrumentation voltage-to-current converter  20  included in an integrated linearization circuit within dashed line  11 . Instrumentation voltage-to-current converter  20  includes two external gain adjustment inputs across which a gain adjusting resistor  23  of resistance R G  is connected to determine the transconductance of instrumentation voltage-to-current converter  20 . 
     Instrumentation voltage-to-current converter  20  produces two essentially equal output currents, including a linearization current I LIN  flowing through conductor  22  and an output current I OUT  which flows out of conductor  21  to a utilization circuit (not shown). Linearization current I LIN  flows from external integrated circuit package lead  22 A through conductor  22 . I LIN  and I OUT  are proportional to the voltage difference between conductors  16  and  18  if the bridge sensor circuit  12  is unbalanced. I LIN  flows from +V CC  into external package lead  22 A through an external resistor  15  of resistance R LIN1 , producing a corresponding voltage on the (+) input of a simple differential amplifier  24 . The (−) input of amplifier  24  is connected to one terminal of an internal resistor  17  of resistance R LIN2 , the other terminal of which is connected to +V CC . The (−) input of amplifier  24  also is connected to the emitter of a PNP transistor  26 . The output of amplifier  24  is connected to the base electrode of PNP transistor  26 . The collector of PNP transistor  26  is connected to conductor  28 , through which a scaled linearization current I IN  flows. By suitably ratioing R LIN1  and R LIN2 , I IN  can be scaled relative to I LIN  to accommodate different ranges of magnitudes of non-linearity errors of bridge sensor circuit  12 . 
     Thus, the current I LIN  generated in the output conductor  22  of instrumentation voltage-to-current converter  20  is scaled according to the ratio of R LIN1  to R LIN2  to produce the scaled linearization current I IN . The direction of I IN  then is either reversed or not reversed, according to the level of polarity control signal V LINPOL , by a current direction switch circuit  30  to produce a bi-directional correction current I CORR . I CORR  then is injected into the voltage divider formed by internal resistors  41  and  42  of resistances R D1  and R D2 , respectively. This voltage divider modulates the initial value V EXCITE(0)  of the excitation voltage V EXCITE  proportionally to the output current I LIN  of instrumentation voltage-to-current converter  20 . Therefore, the value of the linearization constant K LIN  of Equation (1) is given by the expression                K   LIN     =           R   LIN1     ×     R   D2           R   G     ×     R   LIN2         .             (     Equation                 2     )                                
     Since R LIN1  and R G  are external to the integrated circuit indicated by dashed line  11 , the linearization constant K LIN  becomes independent of variations in the values of the resistors R D1 , RD 2 , and R LIN2  formed in an integrated circuit including the circuitry shown within dashed line  11 . The accuracy K LIN  therefore is determined by the matching of R LIN2  and R D2 . 
     The current direction switch circuit  30  allows the user to conveniently set the polarity sign in Eq. 1 above to compensate both positive and negative “bows” of the non-linearity of bridge sensor circuit  12 . Specifically, I IN  flows into current direction switch circuit  30  to produce correction current I CORR  so it flows either into or out of a conductor  32  of a feedback circuit R D2 ,  60 ,  61  (subsequently described) of a differential amplifier  34 , the output of which produces V EXCITE . The direction of I CORR  is controlled by the signal V LINPOL  on external package lead  31 . If V LINPOL  is connected to +V CC , I CORR  flows into conductor  32 , which is connected to the junction between resistors R D1  and R D2 . Conductor  32  also is coupled (for example, as shown in FIG. 1A) to the (−) input of differential amplifier  34 , which also has an input connected to a band gap reference voltage circuit  36 . The correction current I CORR  flowing into the feedback circuit of amplifier  34  causes a modulation of V EXCITE  according to the magnitude and polarity of I CORR . If V LINPOL  is at +V CC  volts, the excitation voltage V EXCITE  on conductor  14  decreases slightly so as to compensate for a non-linearity in bridge sensor circuit  12 . If the signal V LINPOL  on external package lead  31  is connected to ground, then the correction current I CORR  flows out of conductor  32 , causing the excitation voltage V EXCITE  on conductor  14  to be increased slightly so as to correct the nonlinearity of bridge sensor  12 . 
     Thus, the excitation voltage of bridge sensor circuit  12  is equal to a band gap voltage produced by band gap reference voltage circuit  36 , but slightly increased or slightly decreased in proportion to the error voltage between bridge sensor circuit output conductors  16  and  18 . 
     FIG. 1A more accurately shows the details of a conventional Browkaw band gap circuit  36  and the connection of its differential output between the (+) and (−) inputs of differential amplifier  34 . Band gap circuit  36  includes an NPN transistor  60  having its emitter connected to the upper terminal of a resistor  62 , the lower terminal of which is connected to ground. The collector of transistor  60  is connected to the (−) input of differential amplifier  34  and to one terminal of a load resistor  64 , the other terminal of which is connected to conductor  14 . The base of transistor  60  is connected to conductor  32  and the base of an NPN transistor  61  having a substantially larger emitter area than transistor  60 . The base of transistor  61  also is connected to conductor  32 . The emitter of transistor  61  is coupled by resistor  63  to the emitter of transistor  60 , so that a voltage difference proportional to absolute temperature is developed across resistors  62  and  63 . The collector of transistor  61  is connected to the (+) input of differential amplifier  34  and to one terminal of load resistor  65 , the other terminal of which is connected to conductor  14 . 
     The details of instrumentation voltage-to-current converter  20  are shown in FIG. 3, wherein bridge sensor circuit output conductor  16  is connected to the non-inverting input of an operational amplifier  33 , the output of which is connected to the base of an NPN transistor  37 . The emitter of transistor  37  is connected to the inverting input of operational amplifier  33  and to one terminal of the external transconductance-setting resistor R G . The other terminal of resistor R G  is connected to the inverting input of an operational amplifier  35  and to the source of a P-channel JFET (junction field effect transistor)  38 . The gate electrode of JFET  38  is connected to the output of operational amplifier  35 . Bridge sensor circuit output conductor  18  is connected to the inverting input of operational amplifier  35 . 
     The collector of transistor  37  is connected to conductor  22 , so linearization current I LIN  flows through conductor  22 , NPN transistor  37 , resistor R G , and JFET  38 . The drain of JFET  38  is connected to conductor  21 , so the output current I OUT , (which is essentially equal to I LIN ) flows out of conductor  21  into a utilization circuit. (If the user desires a voltage output rather than a current output from the circuit  10  of FIG. 1, a conventional current-to-voltage converter circuit  55  can be coupled as indicated by dashed line  21 A in FIG. 3 to conductor  21  to convert I OUT  to an output voltage V OUT  that represents the analog quantity sensed by bridge sensor circuit  12 .) The configuration of instrumentation voltage-to-current converter  20  shown in FIG. 3 is conventional, being similar to the instrumentation amplifier in the assignee&#39;s XTR105 bridge linearization circuit. 
     The details of one implementation of current direction switch circuit  30  of FIG. 1 are shown in FIG.  2 . In current direction switch circuit  30 , if the two JFETs J 1S  and J 2S  are turned on by setting V LINPOL  to ground, diode-connected transistor Q 4S  is turned off and I IN  flows into the collector of transistor Q 1S , and therefore is mirrored into the collector of transistor Q 2S . Therefore, the correction current I CORR , in effect, flows from conductor  32  into the collector of transistor Q 2S . However, if the value of V LINPOL  is set to +V CC , this turns JFETs J 1S  and J 2S  off, and I IN  flows through diode-connected transistor Q 4S , so I CORR  flows from the emitter of Q 4S  into conductor  32 . 
     FIG. 4 shows a CMOS implementation  30 A of the current direction switch circuit  30  of FIG.  1 . If V LINPOL  is set at +V CC  volts, the linearization current I IN  in conductor  28  flows through a CMOS transmission gate  47  including N-channel MOSFET  47 A and P-channel MOSFET  47 B both connected between conductors  28  and  32  to generate the correction current I CORR , which flows out of terminal  32 . This is because CMOS transmission gate  47  is turned on as a result of setting V LINPOL  at +V CC  volts. A CMOS transmission gate  48 , including N-channel MOSFET  48 A and P-channel MOSFET  48 B both connected between conductors  28  and  50 , is turned off, and a second CMOS transmission gate  51  (including P-channel MOSFET  51 A and N-channel MOSFET  51 B both connected between conductor  50  and ground) is turned on. An N-channel MOSFET  49  is connected between conductor  50  and ground. The drain and gate of MOSFET  49  are connected to conductor  50 , which is also connected to the gate of an N-channel MOSFET  52  having its drain connected to conductor  32  and its source connected to ground. 
     In the circuit of FIG. 4, the logical complement of V LINPOL  needed to operate CMOS transmission gates  47 ,  48  and  51  is generated on conductor  46  by a CMOS inverter  45  having its input connected to conductor  31 . Thus, V LINPOL  is applied to the gate electrodes of N-channel MOSFET  47  A, P-channel MOSFET  482 , and N-channel MOSFET  51 B, and the logical complement of V LINPOL  is applied by conductor  46  to the gates of P-channel MOSFET  47 B, N-channel MOSFET  48 A, and P-channel MOSFET  51 A. 
     If V LINPOL  is set to ground volts in the circuit of FIG. 4, transmission gates  47  and  51  are turned off and transmission gate  48  is turned on. This causes N-channel current mirror MOSFET  52  to be turned on and to therefore conduct a current which is a “mirrored” replication of the scaled linearization current I IN  This mirrored current becomes the correction current I CORR , flowing into conductor  32  through N-channel MOSFET  52  to ground. 
     One advantage of the circuit of FIG. 1 is that only one external lead of the integrated circuit  11  incorporating the linearization circuit is required to set the polarity of the correction current I CORR , that lead being connected to conductor  31 . The amount of linearization can be scaled by selecting/adjusting the resistance of external resistor R LIN1 . 
     Another advantage of the circuit of FIG. 1 is that since R LIN1  is external, its value is independent of on-chip variation in the values of the other resistors, all of which are included on the integrated circuit chip  11 . Note that in Equation (2), R LIN1  and R G  both are external, and R D2  and R LIN2  both are on-chip and therefore have the same variation with processing parameters, etc. 
     A important advantage of the circuit shown in FIG. 1 is that no separate external instrumentation amplifier need be provided by the user, whereas in most applications of a bridge linearization circuit a separate external instrumentation amplifier must be supplied by the user to amplify the bridge output voltage into a useful signal useful for a utilization circuit. Instrumentation amplifiers generally are complex, expensive precision circuits. Furthermore, prior linearization schemes for bridge sensors typically have also required two available leads to adjust the gain of the linearization circuit and another two available leads to set the direction of the correction current to alter the polarity of the modulation of the bridge excitation voltage. The above described linearization circuit provides an economical, single-chip linearization solution wherein no separate instrumentation amplifier is required, and wherein the user needs only to (1) select a suitable value of external resistor R LIN1  to scale the linearization current I LIN , and (2) supply a suitable logical level for V LINPOL  to set the polarity of the correction. Furthermore, only one external lead of the linearization circuit is needed for setting the magnitude of the correction and only one other external lead is needed for setting the polarity of the correction. 
     By using the output signal of the instrumentation voltage-to-current converter  20  in the form of a linearization current for correcting the bridge excitation voltage V EXCITE , and by using simple differential amplifier with an external precision resistor R LIN1  connected between +V CC  and external package terminal  22 A, and by providing a current direction switch circuit  30  requiring only one external package lead  31  to control the polarity of the internal correction current I CORR  and hence the polarity of the modulation of V EXCITE , the total amount of required circuitry is reduced, because no external instrumentation amplifier is needed. 
     Thus, the described bridge sensor linearization circuit provides the user with superior precision, lower product cost, and a smaller package than the prior art, and allows the user to avoid the need for and cost of providing “build-your-own” linearization circuits, which has been a common practice. 
     While the invention has been described with reference to several particular embodiments thereof, those skilled in the art will be able to make the various modifications to the described embodiments of the invention without departing from the true spirit and scope of the invention. It is intended that all elements or steps which are insubstantially different or perform substantially the same function in substantially the same way to achieve the same result as what is claimed are within the scope of the invention. 
     For example, the scaled linearization current I IN  can be produced by a conventional MDAC (multiplying digital-to-analog converter) as indicated by numeral  57  in FIG. 6. A microprocessor or microcontroller circuit  58  could be programmed to provide digital control data  59  to control the scaling of I IN  (relative to I LIN ) by MDAC  57 . Microprocessor circuit  58  also could produce the polarity control signal V LINPOL  on conductor  31  of current direction switch circuit  30 . The technique of FIG. 6 would be well suited to use with a CMOS implementation of the linearization circuit as shown in FIG.  4 .