Abstract:
A method and apparatus for conversion of high voltage AC to low voltage high current DC without using high voltage capacitors or transformers. A single switch is used to perform both the functions of pre-regulation and switching conversion. An input voltage detector determines when the input power AC is below a predetermined voltage limit. A threshold voltage generator provides a threshold voltage corresponding to the output voltage. A voltage comparator coupled to the input voltage detector and threshold voltage generator enables a pulse generator to activate the switch to gate a number of pulses of the input power below the predetermined voltage limit at predetermined frequency to a transformer. The converter regulates its output voltage by changing the input voltage threshold at which it starts switching, instead of using PWM or other known regulation technique.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
     This application claims the benefit under 35 U.S.C. §119(e) of U.S. Provisional Pat. App No. 61/318,600 filed Mar. 29, 2010, and entitled “Switching Regulator,” which is incorporated herein by reference as if set forth in its entirety. 
    
    
     TECHNICAL FIELD 
     The present invention relates generally to the conversion of high voltage alternating current (AC) to low voltage direct current (DC), and more particularly to devices and methods for converting high voltage AC to low voltage high current DC without using high voltage filter capacitors and high voltage switching transformers. 
     BACKGROUND 
     Many applications, such as computer power supply and power supplies in TV and Video sets, require low voltage DC output power for use by analog and digital circuitry. However, the power available to them is the mains power which is high voltage AC, supplied by an AC electric power utility and usually within the range of 80 VAC and 600 VAC. As the mains power is the only power available for use with these types of applications the high voltage AC mains power requires to be converted to low voltage DC power before supplying to the components. 
     The available power supply systems, to provide the high voltage AC to low voltage DC conversion, can be broadly classified into four categories: the mains frequency transformer approach, the high voltage linear regulator approach, the high voltage capacitive coupling approach, and the switching power supply approach. 
     The transformer-based power supplies approach uses a step down mains frequency transformer and some type of wave rectification. These power supplies are isolated from the mains power supply but this isolation requires a bulky and expensive transformer. Further, size of other components, such as capacitors, that are used in conjunction also increases due to the low frequency of operation (50/100 Hz or 60/120 Hz). 
     The high voltage linear regulator approach eliminates the large, costly step down mains frequency transformer, but has the disadvantage of large capacitors (due to the low frequency of 50/100 Hz or 60/120 Hz) and high power dissipation requirements because the excess voltage has to be dropped across the linear pass element. 
     The high voltage capacitive coupling power supplies approach also eliminates the step down transformer and has better efficiency than the high voltage linear regulator approach but has poor regulation and requires large high voltage capacitive elements. 
     The available switching power supplies approach can be further classified into three classes. In the first class are the conventional switching power supplies that can step down high voltage AC from mains power supply to low voltage DC with a very small transformer because of the high switching frequency. These power supplies are also isolated from the mains but the transformer and switch element must be able to withstand the mains voltage and switching transients. Further, the filter capacitors at the input to these switching power supplies must be rated to withstand the maximum peak line voltage and are required to have enough capacitance to maintain the voltage ripple within acceptable limits at the minimum line voltage. These two conditions result in physically large capacitors. These requirements increase the cost and size, though not to the level of the linear power supplies, and make it difficult to use in space-constrained applications, such as telemetry modules for smart electric utility meters, computers, and TV sets. 
     For example,  FIG. 1  is a diagram of a conventional switching power supply used to convert the AC line voltage  110  and produce DC output voltage  170 . The power supply includes a bridge rectifier  120  and a DC-DC converter  100 . It will be understood by those skilled in the art that the filter capacitor  130 , the switch  140 , and the transformer  150  all must be rated to withstand the peak of the maximum input voltage  110  with an adequate margin of safety. For example, for 600 VAC input (480 VAC with 25% safety margin) the rating is 848.5V. Thus, the filter capacitor  130 , the switch  140 , and the transformer  150  must be capable of withstanding 848.5V plus any switching transients that may be generated. 
     In the second class are the switching power supplies that produce low voltage DC from high voltage AC supplied from mains power supply by using a switch that turns on when the input voltage is below the desired output voltage and turns off when this threshold is exceeded. These are now commercially available as single chip solutions with an external switch.  FIG. 2  illustrates such a switching power supply which rectifies and regulates high voltage alternating current without the use of transformers, large capacitive coupling circuits, or high voltage linear regulators. The device includes a rectifier  220 , a control circuit  230  for sensing the output voltage of the rectifier  220  and switching on and off the output of rectifier  220 , a first storage capacitor  240 , a low voltage linear regulator  250  and a second storage capacitor  260 . The control circuit  230  effectively divides the device into a high voltage subsystem  200  and a low voltage subsystem  280 . Although these devices provide advantages in terms of low cost and smaller size, the disadvantages are that they are not isolated from the mains power supply and the linear regulator drastically reduces the efficiency if there is any significant difference between the output voltage of the control circuit  230  and the final output voltage  270 . 
       FIG. 3A  through  FIG. 3D  illustrate a voltage waveform at different points in the circuit of  FIG. 2 . As shown in  FIG. 3A , the voltage waveform  310  of the output of the rectifier  220  to the control circuit  230 , is a rectified form of the input voltage  210  at the same magnitude as the input voltage  210 . The typical output from control circuit  230  for such an input from the rectifier  220  would be the voltage waveform  320  as shown in  FIG. 3B , in which the circuit is closed whenever the full wave rectified voltage is below a prescribed threshold voltage  300 , for example 40 Volts. However, the waveform  330  in  FIG. 3C  shows how the output of the control circuit  230  is altered due to the presence of capacitor  240  in the circuit design of  FIG. 2 . The low voltage linear regulator  250  of  FIG. 2  then produces the regulated DC output voltage waveform  340  as shown in  FIG. 3D , though at a limited output power as noted above. 
     In the third class are the switching power supplies that are a combination of switching power supplies of first and second classes. These use the switching power supply of second class as a pre-regulator for the switching power supply of first class. This results in a power supply that is low cost and compact and is isolated from the high voltage AC mains power supply, but needs two separate switches. The first switch is a high voltage low frequency switch and it acts as a pre-regulator to a second low voltage high frequency switch that does the DC-DC conversion. The second switch may be part of an off the shelf “Brick” DC-DC converter. 
     Such a device is shown in  FIG. 4 , and includes a rectifier  420  for receiving a high voltage AC line power input and for outputting a full wave rectified, high voltage DC, a gating component  430  coupled to the rectifier  420  for receiving the high voltage full wave rectified DC output, acting as the high voltage low frequency switch and outputting an intermediate voltage DC capped by a preset voltage threshold, a first capacitor  440  to smooth out AC ripples, a DC-DC converter  450  coupled to the gating component  430 , for receiving the intermediate voltage DC output, through the first output capacitor  440 , wherein the DC-DC converter  450  is configured to step down the intermediate voltage DC to a desired high current, low voltage DC output using the second low voltage high frequency switch  460  integrated into the DC-DC converter  450  and a second capacitor  470  coupled to the output of the DC-DC converter  450  to further smooth out the high current, low voltage DC output. 
       FIG. 5A  through  FIG. 5D  illustrate a voltage waveform at different points in the circuit of  FIG. 4 . As shown in  FIG. 5A , the bridge rectifier  420  rectifies the AC input voltage  410 , which may range from 80 to 600 VAC, and provides the full wave rectified DC waveform  510 . Now, the gating component  430  turns on at zero crossing and turns off when the full wave rectified DC voltage exceeds a preset voltage threshold V T  (shown as threshold  500  in  FIG. 5A through 5D ), allowing an intermediate DC voltage. Next, the capacitor  440  reduces the AC ripples from the intermediate DC voltage and provides a pre-regulated intermediate DC voltage  530  to the DC-DC converter  450 , including switch  460  and the transformer, as shown in  FIG. 4 . These components step down the pre-regulated intermediate voltage DC  530 , with another capacitor  470  to further reduce the AC ripples, to a predetermined final DC voltage  540 , as shown by curve  540  in  FIG. 5D . 
     The third type of switching power supply is an improvement over the second type because it replaces the linear regulator  250  of  FIG. 2  with a DC-DC converter  450  in  FIG. 4 , and thus improves the current output capability and efficiency. The need for two separate switching elements (high voltage, low frequency switch used for gating and low voltage high frequency switch used for DC-DC conversion) is a disadvantage because it adds cost and complexity. 
     There is therefore a need for improved systems, devices, and circuit designs for converting high voltage AC to low voltage DC without the use of large high voltage filter capacitors or large high voltage switching power supplies or multiple switches, while also providing for high current low voltage DC outputs. Further, there is a need to provide methods, systems, circuit designs, and devices to reduce the size and cost of a power supply module. 
     BRIEF SUMMARY OF THE INVENTION 
     The present invention provides a solution to the above mentioned problems by improving upon the advantages of the switching power supplies of third class, by using a single switch to perform both the functions of pre-regulation and switching conversion and eliminating the need for bulky, high voltage input capacitor. Further, the DC-DC converter of the present invention regulates its output voltage by changing the input voltage threshold below which it starts switching, instead of using pulse width modulation (PWM) or other known regulation technique. 
     Briefly described, aspects of the present invention relate to apparatus and methods for conversion of high voltage AC to low voltage high current DC without using high voltage capacitors or high voltage DC-DC transformers. A single electronically actuated switch is used to perform both the functions of pre-regulation and switching conversion by switching a rectified input power voltage to a transformer and filter capacitor only during such times as the input power voltage is below a predetermined voltage limit and the output power voltage is below a required output DC voltage level. An input voltage detector determines when the input power AC is below a predetermined voltage limit. A threshold voltage generator provides a threshold voltage corresponding to the DC output voltage. A voltage comparator coupled to the input voltage detector and threshold voltage generator enable a pulse generator to activate the switch to gate a number of pulses of the input power, while below the predetermined voltage limit, at predetermined frequency to a transformer. The converter regulates its output voltage by changing the input voltage threshold below which it starts switching, instead of using PWM or other known regulation technique. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Many aspects of the invention can be better understood with reference to the following drawings. The components in the drawings are not necessarily to scale, emphasis instead being placed upon clearly illustrating the principles of the present invention. Moreover, in the drawings, like reference numerals designate corresponding parts throughout the several views. 
         FIG. 1  is a schematic of a prior art conventional switching power supply; 
         FIG. 2  is a schematic of a prior art switching power supply that produces low voltage DC from high voltage AC supplied from mains power supply by using a switch that turns on when the input voltage is below the desired output voltage and turns off when this threshold is exceeded using a control circuit to divide high voltage and low voltage subsystems; 
         FIGS. 3A-3D  illustrate the voltage waveforms corresponding to various locations on the schematic of the prior art switching power supply of  FIG. 2 ; 
         FIG. 4  is a schematic of a prior art switching power supply, using a DC-DC converter and one additional switch; 
         FIGS. 5A-5D  illustrate the voltage waveforms corresponding to various locations on the schematic of the prior art switching power supply of  FIG. 4 ; 
         FIG. 6  is a schematic of a prior art circuit that shows the control scheme of the power supply module of  FIG. 1 , having a pulse width modulator (PWM) for implementing various forms of pulse modulation including off-time modulation, on-time modulation or any combination thereof. 
         FIG. 7  illustrates an embodiment of a preferred power supply module constructed in accordance with aspects of the present invention. 
         FIGS. 8A-8F  illustrate voltage waveforms corresponding to various locations on the schematic of the preferred power supply module of  FIG. 7 . 
         FIG. 9  is a flow chart illustrating the steps of a method taken to reduce high voltage low current AC to low voltage high current DC using a module in accordance with that shown in  FIG. 7 . 
         FIG. 10  illustrates an alternate embodiment of the preferred power supply module according to another aspect of the invention. 
         FIG. 11  is a flow chart illustrating steps of a method taken to reduce high voltage low current AC to low voltage high current DC using a module similar to that shown in  FIG. 10 . 
         FIGS. 12A to 12F  illustrate the voltage waveforms corresponding to various locations on the schematic of  FIG. 10 . 
     
    
    
     DETAILED DESCRIPTION 
     Reference is now made in detail to the description of the embodiments of systems and methods for conversion of high voltage alternating current (AC) to low voltage direct current (DC), as illustrated in the drawings. The invention may, however, be embodied in many different forms and should not be construed as limited to the embodiments set forth herein; rather, these embodiments are intended to convey the scope of the inventions to those skilled in the art. Furthermore, all “examples” given herein are intended to be non-limiting. 
     Various embodiments are described with reference to the drawings, wherein like reference numerals are used to refer to like elements throughout. In the following description, for purposes of explanation, numerous specific details are set forth in order to provide a thorough understanding of one or more embodiments. It may be evident, however, that such embodiments may be practiced without these specific details. In other instances, well-known structures and devices are shown in block diagram form in order to facilitate describing one or more embodiments. 
     Referring to  FIGS. 7 and 8 ,  FIG. 7  is a schematic illustrating a preferred embodiment of a switching power supply module  700  for converting high voltage alternating current (AC) V i  to low voltage direct current (DC) V f  without the need for large high voltage filtering capacitors or high voltage switching power supplies or two separate switches.  FIG. 8A  through  FIG. 8F  illustrate a voltage waveform at different points in the switching power supply module  700 , as will be described in greater detail herein. 
     As shown in  FIG. 7 , a bridge rectifier  709  rectifies the AC input V i , which may range from 80 to 600 VAC, and provides a full wave rectified DC waveform  710  as input to the high voltage switch  711 . In one embodiment, the switch  711  is a Emitter Switched Bipolar Transistor or ESBT manufactured by ST Microelectronics. In an embodiment, the switch  711  starts switching as soon as the input voltage  710  falls below a preset voltage threshold V threshold  (shown as threshold  705  in  FIG. 8 ). Waveform  710  shows the voltage waveform present at the input of the switch  711  when it is switching during the periods indicated by  730  in  FIG. 8B . 
     A pulse generator  714  provides a gating signal on line  735  to the switch  711  at a predetermined frequency, greater than the frequency of the AC input power. Preferably, the frequency of the gating signal, and thus of the output of the switch  711 , is between about 20 kHz and about 100 kHz. In one embodiment, the pulse generator is a TS555IDT low power CMOS timer manufactured by ST Microelectronics. 
       FIG. 8B  also shows the periods  730  during which the switch  711  keeps switching, that is, during the time that the input voltage V rectified    710  remains below V threshold    705 . The waveform  725  of  FIG. 8C  shows the output of the voltage comparator  716  which is high whenever the rectified input voltage  710  is below the threshold voltage  705 . The waveform  735  of  FIG. 8D  shows the output of the pulse generator  714 . The pulse generator  714  generates a train of pulses as long as the output of the voltage comparator  716  remains high and does not generate any pulses when the output of the voltage comparator  716  is low. In one embodiment, the comparator  716  is a TLV3401IDBVR voltage comparator manufactured by Texas Instruments. 
     A threshold voltage generator  718  receives the output voltage V f  and compares this voltage to a predetermined reference voltage V required , which corresponds to the desired DC output voltage level. In one embodiment, the threshold voltage generator is a CA3140 operational amplifier manufactured by Intersil coupled to V f  by a 4N35SR2M optocoupler manufactured by Fairchild Semiconductors and also coupled to a zener-based voltage reference that provides the reference voltage V required . 
     The waveform  745  of  FIG. 8E  shows the amplitude variations of the current pulses delivered by switch  711  to the transformer  712 . It can be seen that the amplitude of the current pulses is proportional to the instantaneous value of the voltage waveform  710 . This fact is used to regulate the output voltage V f  as follows: 
     If V f  falls below the required value V required , the threshold voltage generator  718  increases the threshold voltage  705 . This increases the peak value of the current pulses delivered into the primary winding of the transformer  712 . 
     The energy transferred to the secondary winding by each pulse in discontinuous conduction mode is given by E=½ LI 2  - - - (Equation 1), where E is the energy in Joules, L is the inductance of the primary winding in Henrys and I is the peak value of the primary current in amperes. 
     The peak primary current I during a pulse in discontinuous mode is given by I=(V×t)/L - - - (Equation 2), where V is the input voltage in volts during the pulse, t is the duration of the pulse in seconds and L is the inductance of the primary in Henrys
 
Combining equations (1) and (2) we get  E=V   2   t   2 /2 L   (Equation 3)
 
     If less energy is transferred to the secondary with each pulse than is taken away by load connected across capacitor  770 , voltage V f  falls. Since the energy transferred to the secondary with each pulse is a function of the input voltage during that pulse as shown by equation (3), the Threshold voltage generator  718  samples the output voltage V f  and keeps increasing the threshold voltage till V f  reaches V required . 
     If V f  rises above the required value V required , the threshold voltage generator  718  decreases the threshold voltage  705 . This decreases the peak value of the current pulses delivered into the primary winding of the transformer  712 . 
     Since less energy is transferred to the secondary with each cycle (as given by Equation (3)), the voltage V f  falls. The Threshold voltage generator samples the output voltage V f  and keeps decreasing the threshold voltage till V f  reaches V required . 
     It is important to keep in mind that this control scheme does not require any modulation of the on time or off time of the pulses, although this may be done to provide an additional level of control. 
     The input voltage detector  715  provides a sample of the input AC voltage to the voltage comparator  716  to be compared against the threshold voltage  705 . 
     Still referring to  FIG. 7 , according to another aspect of the invention, a rectifier  713  is coupled between the secondary of the transformer  712  and the capacitor  770  to provide further rectification of the output voltage from the transformer  712 . 
     Further,  FIG. 8F  shows the final (No load) rectified DC voltage output  740  from the rectifier  713  that is connected to the secondary of transformer  712 , followed by the capacitor  770  to smooth out the output  740 . 
     As can be seen, the output waveform under load shows small high frequency (equal to the switching frequency of the switch  711 ) ripples  750  superimposed on a larger low frequency ripple  760 . The high frequency ripple  750  is caused by the switching frequency of the switch  711 , while the low frequency ripple  760  is caused by the off-time (when input voltage is above the threshold and switching is stopped) alternating with the on-time (when the input voltage is below the threshold and the switch  711  is switching). 
     As a result of the above, the transformer  712  and the capacitor  770  never see the full input voltage  710  as long as V threshold  is kept lower than V i . Since voltage for these components is limited, the large (and bulky) high voltage transformers and capacitors that require large portions of printed circuit board (PCB) space are not required in implementations according to the preferred embodiments of the present invention. In an embodiment of the invention for operation with conventional (household and industrial) 120 VAC input power, the voltage rating for the transformer  712  can be as low as 72 volts, and the voltage rating for the capacitor  770  can be as low as 72 volts. It will be appreciated that such low voltage ratings for these components allow an AC-DC power converter constructed as described herein to be compact and low cost due to the ability to employ low voltage rated components. 
     It will also be appreciated that the circuit as described herein has a failure mode that prevents high input voltage from damaging the low voltage rated components. As will be understood, the pulse generator only activates the switch  711  with its pulses so long as the input voltage is below the threshold voltage as determined by the threshold voltage generator. If the input voltage suffers a temporary high voltage spike or overvoltage condition, no pulses will be generated by the pulse generator  714  and thereby prevent high voltage from being coupled to the transformer  712  or filter capacitor  770 . 
       FIG. 9  is a flow chart illustrating the steps of a method or process  900  taken to reduce high voltage low current AC to low voltage high current DC as described before using the module and waveform shown in  FIGS. 7 and 8 , respectively. At step  910 , high voltage AC is obtained from the mains power supply V i  and is rectified to a high voltage DC. At step  912 , a preset threshold voltage V threshold  is determined, such that the switch  711  stops switching when the rectified voltage is above V threshold . At step  914 , a determination is made that whether the rectified voltage V rectified  is above V threshold  or below V threshold . If V rectified  is below V threshold  the process moves to step  916  and the switch  711  keeps switching. However, if V rectified  is above V threshold  then step  918  is executed and the switch  711  stops switching. 
     On determination that V rectified  is below V threshold , at step  920  the rectified DC pulses are provided to the transformer  712 . At step  922  the transformer output pulses are rectified by rectifier  713  to a final low voltage DC. Next, at step  924  AC ripples are smoothed out using the capacitor  770  to produce the final low voltage smooth DC output which is provided to the required components. 
     To control irregularities in the required output voltage due to fluctuations in input voltage, most of the available switching power supplies use a form of output voltage regulation known as Pulse Width Modulation (PWM) to ensure a steady supply to the components. As per PWM, a feedback loop is used to correct the output voltage by changing the on-time or off time of the switching element in the converter. In an embodiment of the present invention, a voltage regulation method has been used, as shown in the  FIG. 9 . As per this, the threshold voltage generator  718  receives the final smoothed out output voltage as a feedback and adjusts the threshold voltage V threshold . Thus, V threshold  is pushed up to a higher voltage if the output voltage falls below the desired value and V threshold  is pulled down to a lower voltage is the output voltage rises above the desired value. This is shown in steps  926  to  932  in the flowchart of  FIG. 9 . 
     As one with skill in the art will appreciate from a closer study of  FIG. 4 , in order to use a standard “off the shelf” DC-DC converter with maximum input voltage capability of a particular voltage, one must set the gating component  430  to “cut-off” at that voltage, i.e. the V threshold . One skilled in the art will further appreciate that the present invention benefits from retaining the feature of the schematic of  FIG. 4  that the transformer  712  and capacitor  770  need not be rated to withstand the full input voltage V i  because they are never exposed to the full input voltage V i  as long as the threshold voltage  705  remains lower than V i . 
     Further, as can be seen from  FIG. 4 , the prior art DC-DC converter  450  keeps running at all times drawing on energy stored in the input capacitor  440 . The gated power supply recharges this capacitor  450  when the input rectified DC is below a preset voltage threshold. Advantageously, as per the present invention, the transformer  712  and switch  711  of the power supply module  700  runs only when the full wave rectified DC is below the voltage threshold V threshold  and no input capacitor is required. 
     Additionally, the switching transformer  712  and the downstream components ( 770 , etc.) never see the full input voltage from the mains power supply V i , and hence are not required to be rated to withstand the full input voltage. These need only be rated to sustain the voltage below the threshold voltage V threshold . There is no need for rating these components according to the line power V i  supplied by the AC power utility, since the high voltages do not propagate beyond the switch  711 . As a result, the switching transformer  712  and the downstream components can be much smaller and more cost effective than a conventional switching power supply. 
     Referring to  FIGS. 10 and 12 ,  FIG. 10  is a schematic illustrating an alternate embodiment of a switching power supply module  1000  for converting high voltage alternating current (AC) to low voltage direct current (DC) without the need for large high voltage filtering capacitors or high voltage switching power supplies or two separate switches.  FIG. 12A  through  FIG. 12F  illustrate a voltage waveform at different points in the switching power supply module  1000 , as will be described in greater detail herein. 
     As shown in  FIG. 10 , a bridge rectifier  709  rectifies the AC input V i , which may range from 80 to 600 VAC, and provides a full wave rectified DC waveform  1010  as input to the high voltage switch  711 . In an embodiment, the switch  711  starts switching as soon as a zero crossing is detected by a zero crossing detector  1014 . The switch  711  stops switching as soon as the enable interval T 1  generated by an Enable interval generator  1025  ends. The waveform of  FIG. 12A  shows the time period T of the full wave rectified waveform  1010 . The waveform of  FIG. 12B  shows the relationship between the time period T and the enable interval T 1  denoted by  1030  in  FIG. 12C   
     The zero crossing detector  1014  triggers the pulse generator control  1015  every time a zero crossing in the input AC waveform V i  is detected. Once triggered, the pulse generator control output  1030  (as shown in  FIG. 12C ) remains high for the duration of the Enable interval. As long as the output of the pulse generator control  1030  is high, the pulse generator  714  keeps outputting pulses  1020  (as shown in  FIG. 12D ) that pulse the switch  711  on and off with each pulse. Each time the switch is pulsed on, a current pulse proportional to the instantaneous value of the rectified AC voltage  1010  is injected into the primary winding of the transformer  712 . The low voltage output pulse at the output of the secondary winding of the transformer  712  is rectified by the rectifier  713  and filtered by the filter capacitor  770  to produce the final output  1040  as shown in  FIG. 12F . The pulse generator is disabled if an overvoltage condition is detected by the optoisolated input voltage detector  715 . 
     The waveform  1015  of  FIG. 12E  shows the amplitude variations of the current pulses delivered by switch  711  to the transformer  712 . It can be seen that the amplitude of the current pulses is proportional to the instantaneous value of the voltage waveform  1010 . This fact is used to regulate the output voltage V f  as follows: 
     If V f  falls below the required value V required , the enable interval generator  1025  increases the length of the enable interval  1030 . This increases the effective input voltage when pulses are being delivered into the primary winding of the transformer  712 . 
     The energy transferred to the secondary winding by each pulse in discontinuous conduction mode is given by E=½V 2 t 2 /L - - - (Equation (3)), where E is the energy in Joules, L is the inductance of the primary winding in Henrys, t is the duration of the pulse in seconds and V is the input voltage in volts during the pulse. 
     Since more energy is transferred to the secondary with each cycle, the voltage V f  rises. The Threshold voltage generator samples the output voltage V f  and keeps increasing the threshold voltage till V f  reaches V required . 
     If V f  rises above the required value V required , the enable interval generator  1025  decreases the length of the enable interval  1030 . This decreases the effective input voltage when current pulses are being delivered into the primary winding of the transformer  712 . 
     Since less energy is transferred to the secondary with each cycle (as given by (3)), the voltage V f  falls. The enable interval generator  1025  samples the output voltage V f  and keeps decreasing the enable interval till V f  reaches V required . 
     It is important to keep in mind that this control scheme does not require any modulation of the on time or off time of the pulses although this may be done to provide an additional level of control. 
     Further,  FIG. 12F  shows the final (No load) rectified DC voltage output  1040  from the rectifier  713  that is connected to the secondary of transformer  712 , followed by the capacitor  770  to smooth out the output  740 . 
     As can be seen, the output waveform under load shows small high frequency (equal to the switching frequency of the switch  711 ) ripples  1050  superimposed on a larger low frequency ripple  1060 . The high frequency ripple  1050  is caused by the switching frequency of the switch  711 , while the low frequency ripple  1060  is caused by the off-time (when  1030 , the output of the pulse generator control  1015  is low and switching is stopped) alternating with the on-time (when  1030 , the output of the pulse generator control  1030  is high and the switch  711  is switching). 
     As a result of the above, the transformer  712  and the capacitor  770  never see the full input voltage  710  as long as the enable time T 1  is kept less than T/4 where T is the time period of the AC input waveform. Since voltage for these components is limited, the large (and bulky) high voltage transformers and capacitors that require large portions of printed circuit board (PCB) space are not required in implementations according to the preferred embodiments of the present invention. 
       FIG. 11  is a flow chart illustrating the steps  1100  taken to reduce high voltage low current AC to low voltage high current DC as described before using the module and waveform shown in  FIGS. 10 and 12 , respectively. At step  1010  high voltage AC is obtained from the mains power supply V i  and is rectified to a high voltage DC. At step  1012 , a preset enable time T 1  is determined, such that the switch  711  stops switching when the time elapsed since the last zero crossing exceeds T 1 . At step  1114 , a determination is made that whether the time elapsed since the last zero crossing is greater than or less than the enable interval T 1 . If t&lt;T 1  or t&gt;(T/2−T 1 ) the process moves to step  1116  and the switch  711  keeps switching. However, if this condition is found to be false, then step  1115  is executed and the switch  711  stops switching. 
     On determination that t&lt;T 1  or t&gt;(T/2−T 1 ), at step  1118  the rectified DC pulses are provided to the transformer  712 . At step  1120  the transformer output pulses are rectified by rectifier  713  to a final low voltage DC. Next, at step  1121  AC ripples are smoothed out using the capacitor  770  to produce the final low voltage smooth DC output which is provided to the required components. 
     To control irregularities in the required output voltage due to fluctuations in input voltage, most of the available switching power supplies use a form of output voltage regulation known as Pulse Width Modulation (PWM) to ensure a steady supply to the components. As per PWM, a feedback loop is used to correct the output voltage by changing the on-time or off time of the switching element in the converter. In this alternate embodiment of the present invention, a voltage regulation method has been used, as shown in the  FIG. 11 . As per this, the enable interval generator  1025  receives the final smoothed out output voltage as a feedback and adjusts the length of the enable interval. Thus, the enable interval is increased (thus increasing the voltage of the voltage pulses) if the output voltage falls below the desired value and decreased (thus decreasing the voltage of the voltage pulses) if the output voltage rises above the desired value. This is shown in steps  1122  to  1130  in the flowchart of  FIG. 11 . 
     As one with skill in the art will appreciate from a closer study of  FIG. 4 , in order to use a standard “off the shelf” DC-DC converter with maximum input voltage capability of a particular voltage, one must set the gating component  430  to “cut-off” at that voltage, i.e. the V threshold . One skilled in the art will further appreciate that the present invention benefits from retaining the feature of the schematic of  FIG. 4  that the transformer  712  and capacitor  770  need not be rated to withstand the full input voltage V i  because they are never exposed to the full input voltage V i  as long as the enable interval  1030  remains shorter than T/4 (one quarter of the time period of the input AC waveform). 
     There is no need for rating these components according to the line power V i  supplied by the AC power utility, since the high voltages do not propagate beyond the switch  711 . As a result, the switching transformer  712  and the downstream components can be much smaller and more cost effective than a conventional switching power supply. 
     The foregoing description of the exemplary embodiments of the invention has been presented only for the purposes of illustration and description and is not intended to be exhaustive or to limit the invention to the precise forms disclosed. Many modifications and variations are possible in light of the above teaching. 
     The embodiments were chosen and described in order to explain the principles of the invention and their practical application so as to enable others skilled in the art to utilize the invention and various embodiments and with various modifications as are suited to the particular use contemplated. Alternative embodiments will become apparent to those skilled in the art to which the present invention pertains without departing from its spirit and scope.