Abstract:
A power supply device includes a first semiconductor switching device for controlling an alternating input current waveform, a smoothing capacitor to which a rectified voltage is applied, and an inverter that converts the rectified voltage into alternating current via a step-up chopper. The step-up chopper includes an inductor and a diode connected between the smoothing capacitor and inverter, and a second semiconductor switching device connected to the inductor and diode. The power supply device further includes an instantaneous voltage drop compensation function whereby the energy of the smoothing capacitor is supplied by an operation of the step-up chopper to the inverter when there is an instantaneous voltage drop in an alternating current power supply voltage. MOSFETs with a breakdown voltage lower than that of the first semiconductor switching device are connected between terminals of the step-up chopper, thus further reducing loss in comparison with when a bypass diode is used.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a power supply device having a function of compensating for an instantaneous voltage drop in an alternating current power supply voltage, and specifically, relates to technology for increasing the efficiency and reducing the size of the device. 
     2. Description of the Background Art 
       FIG. 3  shows first heretofore known technology of a power supply device having a function of compensating for an instantaneous voltage drop in an alternating current power supply voltage. 
     In  FIG. 3 , reference sign  1  is an alternating current power supply, and a rectifier circuit formed of diodes  2  to  5  is connected to both ends of the alternating current power supply. An inductor  6  and a semiconductor switching device  7 , such as a MOSFET, are connected in series to either end of a series circuit of the diodes  4  and  5 , and a diode  8  and smoothing capacitor  9  are connected in series to either end of the switching device  7 . 
     Direct current input terminals of an inverter INV formed of semiconductor switching devices  10  to  13  are connected to either end of the smoothing capacitor  9 , and a primary winding  14   a  of a transformer  14  is connected to alternating current output terminals of the inverter INV. 
     A rectifier circuit formed of diodes  15  to  18  is connected to both ends of a secondary winding  14   b  of the transformer  14 . An inductor  19  and a capacitor  20  are connected in series to either end of a series circuit of the diodes  17  and  18 , and a load  21  is connected to both ends of the capacitor  20 . 
     Functions required of the above-described power supply device are as follows: 
     converting an alternating current input voltage to a direct current voltage of a desired size, and keeping output voltage constant regardless of fluctuation in input voltage and load current; 
     isolating an alternating current input unit and a direct current output unit; and 
     controlling an alternating input current to a sinusoidal wave with a power factor of practically 1. 
     Furthermore, when the load  21  is one of which reliability is required, as with an information and communication instrument, the power supply device is required to have a function whereby it is possible to maintain a constant output voltage even in the event of a drop in the voltage of the alternating current power supply  1  for a period of in the region of a few milliseconds to a few cycles, a so-called instantaneous voltage drop (this function will hereafter be referred to as an instantaneous voltage drop compensation function). 
     A description will be given, while referring to  FIGS. 4A to 4E , of operations for realizing these functions. 
     In  FIG. 4A , an input voltage V in  from the alternating current power supply  1  has a sinusoidal waveform, while the output voltage (rectified voltage) of the rectifier circuit formed of the diodes  2  to  5  has the waveform indicated by V r1  in  FIG. 4B . Herein, on the switching device  7  of  FIG. 3  being turned on when V in  has, for example, positive polarity, current flows along a path from the alternating current power supply  1  through the diode  2 , inductor  6 , switching device  7 , and diode  5  to the alternating current power supply  1 , the rectified voltage V r1  is applied across the inductor  6 , and a current I L  increases. 
     Also, on the switching device  7  being turned off, the current I L  flows along a path from the alternating current power supply  1  through the diode  2 , inductor  6 , diode  8 , smoothing capacitor  9 , and diode  5  to the alternating current power supply  1 . At this time, a voltage equivalent to the difference between a voltage E d1  of the smoothing capacitor  9  and the input voltage V in  is applied to the inductor  6 , but as E d1  is kept higher than a peak value of V in  by an operation of the circuit, the current I L  decreases. 
     It is possible to control the waveform and amplitude of the current I L  by controlling the duty ratio of the turning on and off of the switching device  7 . When the waveform of the current I L  is the kind of sinusoidal rectified waveform in  FIG. 4B  (ripple is ignored for the sake of simplification), an input current I in  has a sinusoidal waveform, as shown in  FIG. 4A . Also, by controlling the amplitude of I L  in accordance with the load power, it is possible to keep the voltage E d1  of the smoothing capacitor  9  constant, and thus possible to keep the output voltage of the inverter INV constant. 
     Herein,  FIGS. 4C to 4E  show operations and voltage waveforms of each portion of the switching devices  10  to  13  when there is and is not an instantaneous voltage drop compensation function. 
     The inverter INV formed of the switching devices  10  to  13  converts the voltage E d1  of the smoothing capacitor  9  into a high frequency alternating current voltage. A positive voltage V t  is applied to the primary winding  14   a  of the transformer  14  on the switching devices  10  and  13  being turned on, while a negative voltage V t  is applied to the primary winding  14   a  on the switching devices  11  and  12  being turned on, as shown in  FIGS. 4C and 4D . By the positive and negative voltages V t  being applied alternately to the primary winding  14   a  in this way, the high frequency alternating current voltage V t  is input into the transformer  14  (the cycles of V in  and V t  are represented as being of the same extent in  FIGS. 4A to 4E  for the sake of easier understanding, but in general, V in  is of a commercial frequency of 50 or 60 Hz, while V t  is of a frequency of a few kilohertz or more in order to reduce the size of the transformer). 
     The transformer  14  isolates and transforms the input high frequency alternating current voltage V t  and, after converting the voltage across the secondary winding  14   b  of the transformer  14  into a rectified voltage V r2  of  FIG. 4E  using the rectifier circuit formed of the diodes  15  to  18 , smoothes the voltage using the inductor  19  and capacitor  20 , and applies the voltage to the load  21  as an output voltage V out . The output voltage V out  can be controlled by the duty ratio (hereafter referred to as the inverter duty ratio) of the turning on of the switching devices  10  and  13  or switching devices  11  and  12 . 
     Operations when the power supply device includes an instantaneous voltage drop compensation function and the alternating current power supply  1  is sound are as shown under “Normal time” in  FIGS. 4C to 4E . 
     As opposed to this, under “Time of instantaneous voltage drop”, the input power decreases due to the occurrence of an instantaneous voltage drop in the voltage of the alternating current power supply  1 , the voltage E d1  drops, and the amplitude of the voltage V t  also decreases. However, provided that the drop in the voltage E d1  is within a predetermined range, it is possible to keep the average value of the voltage V t  constant by increasing the duty ratio of the turning on of the switching devices  10  and  13  or switching devices  11  and  12 , and to maintain the predetermined rectified voltage V r2  and, by extension, the output voltage V out . 
     However, when an instantaneous voltage drop compensation function is provided, the efficiency of the device decreases. This is for the following reasons. 
     In order to maintain the constant output voltage V out  even when the voltage E d1  has dropped to a certain extent, it is necessary that the transformation ratio of the transformer  14  (the value of n in n:1, which is the turn ratio between the primary winding  14   a  and secondary winding  14   b ) is smaller than an essentially known optimum value. For example, when the voltage E d1  is kept constant at 400V and the output voltage V out  is 10V at a normal time, the transformation ratio of the transformer  14  necessary to operate the inverter INV at a maximum duty ratio is 400:10, that is, n=40 (for the sake of simplification, voltage drop in the circuit is ignored here). 
     Meanwhile, the transformation ratio necessary in order to maintain the output voltage V out  at 10V even when the voltage E d1  drops as far as 200V is 200:10, that is n=20. When setting the transformation ratio n under this condition, operation is carried out with an inverter duty ratio of approximately 0.5 in order to maintain V out  at 10V at a normal time, when the voltage E d1  is 400V. 
     In this case, the amplitude of current flowing on the primary side of the transformer  14  is 1/n that of current flowing through the inductor  19 , but when the power supply device has an instantaneous voltage drop compensation function, the transformation ratio n, whose original optimum value is 40, becomes 20, and the value of current flowing on the primary side of the transformer  14  increases due to the providing of the instantaneous voltage drop compensation function. Because of this, loss occurring in the switching devices  10  to  13  and primary winding  14   a  increases. 
     The rectified voltage V r2  is approximately E d1 /n, but as the voltage applied to the diodes  15  to  18  at a normal time increases when the transformation ratio n decreases, it is necessary to use parts with a high breakdown voltage as the diodes  15  to  18 . Generally, there is a tendency for loss in a semiconductor part to increase under the same conditions the higher the breakdown voltage, because of which loss occurring in the device increases. 
     Also, at a normal time and when there is an instantaneous voltage drop compensation function, a period for which the voltage V r2  is not applied lengthens, because of which the value of the inductance of the inductor  19  necessary in order to smooth the voltage V r2  increases. In the case in which there is no instantaneous voltage drop compensation function in  FIGS. 4D and 4E , the rectified voltage V r2  is not applied for the very short time in which the polarity of the voltage V t  switches in the period during which the rectified voltage V r2  drops from a predetermined value to 0V, but at a normal time and when there is an instantaneous voltage drop compensation function, the rectified voltage V r2  is not applied for a period equivalent to one-half of one cycle, and for this period it is necessary that energy is supplied to the load  21  by the inductor  19 . 
     For these reasons, the inductor  19  increases in size, which results in an increase in the overall size of the device and an increase in loss occurring in the inductor  19 . 
     The circuit shown in  FIG. 5  is known as second heretofore known technology whereby the above-described kind of increase in loss from the inverter INV onward is avoided. 
     In  FIG. 5 , an inductor  22  and a diode  24  are connected in series to a positive side direct current bus between the smoothing capacitor  9  and inverter INV, and a semiconductor switching device  23  is connected between a connection point of the inductor  22  and diode  24  and a negative side direct current bus. Also, a smoothing capacitor  25  is connected between the cathode of the diode  24  and the negative side direct current bus. A step-up chopper is configured of the inductor  22 , diode  24 , switching device  23 , and smoothing capacitor  25 . As the other configurations in  FIG. 5  are the same as in  FIG. 3 , a description thereof will be omitted. 
     It is possible to control current flowing through the inductor  22  with the step-up chopper, using the same kind of operation as that of the circuit formed of the inductor  6 , switching device  7 , and diode  8 , as a result of which it is possible to obtain an output voltage E d2  higher than the input voltage E d1 . That is, using an operation of the step-up chopper, it is possible to keep the voltage E d2  of the capacitor  25  constant (for example, 400V) even when the voltage E d1  of the capacitor  9  drops (for example, from 400V to 200V) when there is an instantaneous voltage drop in voltage, and a design wherein the transformation ratio is n=40 is possible in the previously described example. 
     A circuit wherein a drop in alternating current power supply voltage is compensated for using the above-described step-up chopper is shown in JP-A-2-241371 (page 2, bottom right column, line 1 to page 3, top left column, line 4,  FIG. 2  and the like). 
     However, the circuit shown in  FIG. 5  has the following drawbacks. 
     That is, the inductor  22 , switching device  23 , and diode  24  configuring the step-up chopper generate loss. While inevitable in a case in which the switching device  23  is carrying out a turning on or turning off operation, loss also occurs when the voltage E d1  is sufficiently high and the switching device  23  is stopped, due to the winding resistance of the inductor  22  and the forward voltage drop of the diode  24 . Because of this loss, the amount by which the circuit loss from the inverter INV onward is reduced is cancelled out. 
     Also, as current is constantly flowing through the inductor  22 , an inductor  22  with a large current capacity is needed, despite the time for which the stepping-up operation is carried out being extremely short. 
     Furthermore, a large capacity smoothing capacitor  25  is needed in order to absorb ripple current generated from the inverter INV. As the one smoothing capacitor  9  has a sufficiently large capacity in order to supply energy when there is an instantaneous voltage drop, the smoothing capacitor  9  can perform the role of absorbing both the ripple current generated by the circuit formed of the inductor  6 , switching device  7 , and diode  8  and the ripple current of the inverter INV in the circuit of  FIG. 3 . 
     However, as the inductor  22  is inserted between the inverter INV and smoothing capacitor  9  in the circuit of  FIG. 5 , a high frequency ripple current can not pass. Because of this, the separate smoothing capacitor  25  is needed, because of which the size of the device increases. 
     A circuit wherein energy when there is an instantaneous voltage drop is supplied by another capacitor charged in advance is shown in JP-A-8-185993 (paragraphs [0023] to [0028],  FIG. 1  and the like). However, as the circuit disclosed in JP-A-8-185993 (paragraphs [0023] to [0028],  FIG. 1  and the like) is also such that a capacitor that absorbs ripple at a normal time and a capacitor that supplies energy when there is an instantaneous voltage drop are separated, it is not possible to avoid an increase in the size of the device. 
     The circuit shown in  FIG. 6  is known as third heretofore known technology whereby the above-described increase in the size of the device is avoided. 
     In  FIG. 6 , reference sign  101  is a bypass diode connected between the cathodes of the diodes  8  and  24 , while the other configurations are the same as in  FIG. 5 . At a normal time when the alternating current power supply  1  is sound, the switching device  23  does not operate, and the step-up chopper formed of the inductor  22 , switching device  23 , diode  24 , and capacitor  25  is bypassed by the diode  101 . Herein, as the voltages E d1  and E d2  are smoothed direct current voltages both at a normal time and when there is a instantaneous voltage drop, the diode  101 , unlike the diode  24 , does not need to be capable of rectifying a high frequency, so it is possible to use a low speed diode. As the forward voltage of a low speed diode is in the region of one-half compared to that of a high speed diode, and no current flows through the inductor  22  at a normal time, it is possible to reduce loss considerably in comparison with that in the circuit of  FIG. 5 . 
     Furthermore, as the time for which the inductor  22  is energized is a few tens of milliseconds or less when there is an instantaneous voltage drop, it is possible to use a short time rated inductor with thin windings as the inductor  22 , and thus possible to reduce the size of the inductor  22  considerably in comparison with that in the circuit of  FIG. 5 . Also, as there is no longer an inductor interposed between the smoothing capacitor  9  and inverter INV owing to the bypassing operation of the diode  101 , it is possible for the smoothing capacitor  9  to absorb the ripple current of the inverter INV too at a normal time. Because of this, it is sufficient that the smoothing capacitor  25  withstands only the ripple of the step-up chopper and inverter INV during an instantaneous voltage drop period, and thus possible to use a capacitor with an extremely small capacity. 
     This method is shown in, for example, JP-A-2010-41910 (paragraphs [0040] to [0046],  FIG. 1  and the like). 
     SUMMARY OF THE INVENTION 
     Meanwhile, in recent years, against a background of global environment problems and the like, ever higher efficiency has been required of power supply devices too. Because of this, thorough loss reduction is seen to be needed, and a need to also reduce loss caused by the bypass diode  101  has newly arisen. 
     To date, a method commonly called synchronous rectification has been known as a method of reducing loss caused by a diode, wherein synchronous rectification is such that the diode is substituted with MOSFETs. It is known that a MOSFET not only causes current to flow in a forward direction, that is, from the drain to the source, when voltage is applied to the gate, but also causes current to flow in a reverse direction, that is, from the source to the drain. As a MOSFET in a conductive condition has a property such that the resistivity, that is, the voltage drop, is proportional to the current, the voltage drop is smaller than that of a diode within a range wherein the conducting current is small with respect to the current capacity. Utilizing this, the diode  101  of  FIG. 6  is substituted with MOSFETs, and it is possible to suppress loss by increasing the number of parallel MOSFETs or current capacity thereof, that is, by reducing the resistance value when conductive (hereafter called the on-state resistance). 
     However, the voltage drop of the diode  101  is already in the region of 1V or less, and when using MOSFETs with the same breakdown voltage and specifications as, for example, the switching device  7  in order to obtain the advantage of reducing the voltage drop sufficiently with respect to the voltage drop of the diode  101 , it is necessary that the number of parallel MOSFETs is equivalent to the number of switching devices  7 , which leads to a rise in the device cost. 
     Therefore, an object of the invention is to provide a power supply device wherein loss caused by a heretofore known bypass diode is reduced without leading to a rise in cost. 
     In order to achieve the object, a power supply device according to a first aspect of the invention includes a rectifier circuit that rectifies an alternating current power supply voltage, a first semiconductor switching device for controlling an input current waveform of the rectifier circuit to a sinusoidal wave, a smoothing capacitor to which the output voltage of the rectifier circuit is applied, and an inverter, into a direct current side of which the voltage across the smoothing capacitor is input via a step-up chopper. The inverter converts a direct current input voltage into alternating current voltage and supplies the alternating current voltage to a load. The step-up chopper includes an inductor and diode mutually connected in series to a positive side direct current bus between the smoothing capacitor and inverter and a second semiconductor switching device connected between a connection point of the inductor and diode and a negative side direct current bus. The power supply device further includes an instantaneous voltage drop compensation function whereby the energy of the smoothing capacitor is supplied by an operation of the step-up chopper to the inverter when there is an instantaneous voltage drop in the alternating current power supply voltage and the output voltage of the inverter is maintained at a constant value. 
     Further, the aspect of the invention is such that a plurality of MOSFETs with a breakdown voltage lower than that of the first semiconductor switching device are connected in series between a connection point of the smoothing capacitor and inductor and a connection point of the diode and inverter. 
     The plurality of MOSFETs can be turned on in a condition wherein the potential difference across the series circuit of the MOSFETs is practically zero. 
     In other words, it can be advantageous when the plurality of MOSFETs are turned on at a time of normal operation other than when there is an instantaneous voltage drop in the alternating current power supply voltage, in a condition wherein the step-up chopper does not operate, and the voltage of the smoothing capacitor and the direct current input voltage of the inverter are practically equal. 
     Also, the plurality of MOSFETs can be driven by the same gate drive circuit, and a diode can be connected between the gate electrode of at least one of the plurality of MOSFETs and the gate drive circuit. 
     A direct current power supply having the potential of the connection point of the smoothing capacitor and inductor as a reference potential may be used as a drive power supply of the gate drive circuit, or alternatively, a capacitor in which a charge is accumulated by a turning on and off of the first semiconductor switching device and a fixed voltage element that keeps the voltage of the capacitor constant may be included instead of the direct current power supply. 
     The power supply device according to the aspect of the invention can be configured as an isolated AC-DC conversion device wherein the alternating current output voltage of the inverter is isolated and converted by a transformer, and the output voltage of the transformer is rectified by a rectifier circuit and supplied to a load. 
     According to the invention, it is possible to realize a highly efficient, small-sized, low cost power supply device by reducing loss caused by a heretofore known bypass diode that bypasses a step-up chopper, and simplifying the drive power supply of a series circuit of MOSFETs connected instead of the bypass diode. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a circuit diagram showing a first embodiment of the invention; 
         FIG. 2  is a circuit diagram showing a second embodiment of the invention; 
         FIG. 3  is a circuit diagram showing first heretofore known technology; 
         FIGS. 4A to 4E  are waveform diagrams showing operations of  FIG. 3 ; 
         FIG. 5  is a circuit diagram showing second heretofore known technology; and 
         FIG. 6  is a circuit diagram showing third heretofore known technology. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Hereafter, a description will be given, based on the drawings, of embodiments of the invention. 
       FIG. 1  is a circuit diagram showing a first embodiment of the invention. The first embodiment is such that the circuit described hereafter is added to the circuit shown in  FIG. 5 . The same reference signs are given to portions the same as those in  FIG. 5 , and a description thereof is omitted. 
     That is, MOSFETs  201  and  202  are connected in series between the cathode of a diode  8  and the cathode of a diode  24 , and resistors  203  and  204  with equal resistance values are connected between the source electrode and gate electrode of the MOSFETs  201  and  202  respectively. Also,  207  is a gate drive circuit, and a direct current power supply  208  is connected to both ends of the gate drive circuit  207 . Agate drive signal G output from the gate drive circuit  207  is input directly into the gate electrode of the one MOSFET  201 , and input via a diode  206  into the gate electrode of the other MOSFET  202 . Furthermore, a resistor  205  is connected between the gate electrode and source electrode of the MOSFET  202 . 
     In the above-described configuration, devices with a breakdown voltage lower than that of a first semiconductor switching device  7 , formed of a MOSFET, are used as the MOSFETs  201  and  202 . In  FIG. 1 , another semiconductor switching device  23 , formed of a MOSFET, is referred to as a second semiconductor switching device. 
     Herein, in a circuit in which a voltage E d1  of a smoothing capacitor  9  is in the region of 400V, a device with a breakdown voltage of 500V or 600V is commonly used as the switching device  7 , but the breakdown voltage of both the MOSFETs  201  and  202  is 100V. The voltage applied to the series circuit of the MOSFETs  201  and  202  is a voltage equivalent to the difference between E d2  and E d1 , meaning in practical terms that the lower limit value of E d1  when the smoothing capacitor  9  discharges on the occurrence of an instantaneous voltage drop is in the region of one-half of E d2 . For example, when E d2  is 400V, the lower limit value of E d1  is in the region of 200V. This is because, as it is necessary to increase the circuit current in inverse proportion to the voltage drop in order to obtain the same power, it is not advisable in terms of part current capacity to cause the smoothing capacitor  9  to discharge until E d1  becomes too low. 
     Under the conditions whereby E d2  is 400V and the lower limit value of E d1  is in the region of 200V, the voltage applied to the series circuit of the MOSFETs  201  and  202  is 200V or less, which can sufficiently accommodate a case wherein two MOSFETs with a breakdown voltage of 100V are connected in series. Herein, the characteristics of MOSFETs with a breakdown voltage of in the region of 100V or less have noticeably improved in recent years, and MOSFETs with an extremely low on-state resistance are commercially available. Because of this, even when a plurality of small MOSFETs with a breakdown voltage of 100V or less are connected in series, it is possible for the series on-state resistance to be considerably lower than the on-state resistance of the switching device  7 . 
     Note that a series connection of switching devices is generally more difficult than a parallel connection. This is because, as the drive potential of each series device is different, it is necessary to individually provide drive circuits with independent potential, which is disadvantageous in terms of size and cost, and also because a voltage unbalance occurs in each device when switching in the event of even a slight difference in timing between the drive circuits, and there is a danger of overvoltage breakdown being caused in a device in which voltage is concentrated. Consequently, it is almost always the case that a series connection of switching devices is carried out unavoidably in a case such as when the breakdown voltage in a high voltage circuit is insufficient with one device, and it is rare that switching devices with low breakdown voltage are deliberately connected in series under conditions whereby it is possible to secure breakdown voltage without connecting in series. 
     In the embodiment under discussion, however, on the voltage E d1  starting to drop on the occurrence of an instantaneous voltage drop, the MOSFETs  201  and  202  are turned off before the switching device  23  starts switching, and the MOSFETs  201  and  202  are turned on again under conditions whereby the instantaneous voltage drop is finished, the voltage E d1  rises again, and the switching device  23  has stopped switching. That is, as switching of the MOSFETs  201  and  202  is carried out only when almost no voltage is applied to the series circuit thereof, there is no need to align the switching timings between the MOSFETs  201  and  202 . Also, as there is no problem with switching loss for the same reason, there is no need for the kind of high speed switching that is carried out within, for example, 1 μs, and it is sufficient that switching is possible in a time, for example, a few microseconds, such that it is possible to respond to a circuit voltage change. 
     Because of this, according to the embodiment, it is possible to configure the drive circuit of the MOSFETs  201  and  202  extremely easily. In  FIG. 1 , the gate drive circuit  207  carries out an operation connecting a G point in the drawing to one of a V point or S point (reference potential points) at either end of the direct current power supply  208 , and carries out a switching of the MOSFET  201  by adjusting the gate-to-source voltage of the MOSFET  201  to the voltage of the direct current power supply  208  or to 0V. Herein, as the MOSFET  201  is turned on and the voltage across the MOSFET  201  becomes extremely low when voltage is applied between the G point and S point, a voltage practically equivalent to the voltage of the direct current power supply  208  is also applied to the gate of the MOSFET  202 , and the MOSFET  202  is also turned on. 
     Meanwhile, when the voltage between the G point and S point is 0V, the MOSFET  201  is turned off, the gate-to-source parasitic capacitance (not shown) of the MOSFET  202  is discharged by the resistor  205 , and the gate-to-source voltage of the MOSFET  202  also presently becomes 0V, because of which the MOSFET  202  is also turned off. 
     When E d2  becomes higher than E d1  owing to a step-up chopper operation, the source potential of the MOSFET  202  becomes higher than the potential of the S point, but as the diode  206  shares the difference in potential from that of the gate drive circuit  207  at this time, excessive reverse voltage is prevented from being applied between the gate and source of the MOSFET  202 . Also, the voltage of the MOSFETs  201  and  202  at this time is divided practically equally between the resistors  203  and  204 . 
     As described above, when taking into consideration the appropriate conditions in the circuit of  FIG. 1 , that is, that switching is carried out in a condition wherein almost no voltage is applied to the series circuit of the MOSFETs  201  and  202  and that high speed is not required in the switching of the MOSFETs  201  and  202 , no problem occurs even when using the MOSFETs  201  and  202  connected in series. 
     The same kind of operation is also possible in a circuit wherein three or more MOSFETs with a breakdown voltage lower than that of the switching device  7  are connected in series, although not shown in the drawing, by the gates of MOSFETs from the second stage onward being connected via a diode to the same gate drive circuit  207 , in the same way as the MOSFET  202  in  FIG. 1 , and the MOSFETs being turned on sequentially from the first stage MOSFET  201 . 
     Next,  FIG. 2  is a circuit diagram showing a second embodiment of the invention. The second embodiment relates to the configuration of the direct current power supply  208  in  FIG. 1 . 
     In  FIG. 2 , a parallel circuit of a Zener diode  305  and capacitor  302  is connected to both ends of the gate drive circuit  207 , and diodes  303  and  304  are connected in series with the polarity shown in the drawing to either end of the parallel circuit. Also, a connection point P 2  of the diodes  303  and  304  is connected to the anode of the diode  8  via a capacitor  301 . 
     As the potential of the S point in  FIG. 1  differs from the source potential (an N point potential) of the switching devices  7  and  23 , it is necessary to drive the MOSFETs  201  and  202  with the direct current power supply  208 , which has a reference potential differing from that of the drive circuits of the switching devices  7  and  23 . However, as the MOSFETs  201  and  202  are substitutes for a diode  101  of  FIG. 6 , which intrinsically has no need of a drive circuit, it is not desirable that the device should increase in size and the cost rise due to the drive power supply, or the like, of the MOSFETs  201  and  202 . 
     Therefore, the second embodiment shown in  FIG. 2  is for easily realizing a MOSFET  201  and  202  drive power supply with a reference potential differing from that of the switching devices  7  and  23 . 
     The MOSFETs  201  and  202  are turned on at a time of steady state operation, and at this time, the switching device  7  carries out switching. The potential of a P 1  point with respect to the N point of  FIG. 2  varies at a high frequency between the voltage E d1  and 0V owing to the switching operation of the switching device  7 . 
     On the switching device  7  being turned on, current flows along a path from the smoothing capacitor  9  through the diode  304 , capacitor  301 , and switching device  7  to the smoothing capacitor  9 , and the capacitor  301  is charged to a voltage practically equivalent to E d1 , for example, 400V. On the switching device  7  being turned off, the potential of the P 1  point rises, and the diode  8  presently becomes conductive, because of which the potential of the P 1  point becomes practically equivalent to the potential of the S point. As the potential of a P 2  point in the drawing becomes higher than that of the S point through this process, current flows along a path from the capacitor  301  through P 2 , the diode  303 , and the capacitor  302  to the S point, and a charge accumulated in the capacitor  301  shifts to the capacitor  302 . 
     The voltage of a capacitor is inversely proportional to the capacitance thereof, because of which, when the capacitance of the capacitor  301  is set to, for example, one-thousandth of that of the capacitor  302 , a charge of 400V accumulated in the capacitor  301  is shifted to the capacitor  302  by one switching of the switching device  7 , and the voltage of the capacitor  302  rises by 0.4V. 
     When the above-described operation is repeated by the high frequency switching of the switching device  7 , the voltage of the capacitor  302  gradually rises, but the power consumed by the gate drive circuit  207  and resistor  205  increases, because of which, in principle, the power supplied from the capacitor  302  and the power consumed by the gate drive circuit  207  and the like become balanced at a certain point, and the voltage of the capacitor  302  becomes constant. In practice, as it is difficult to manage the voltage at the point of balance, the voltage of the capacitor  302  is limited to a constant value by the Zener diode  305  connected in parallel to the capacitor  302 . 
     As is well known, hardly any power is consumed at the gate while a MOSFET is maintained in an on-state, because of which, it is possible to use small parts with an extremely small current capacity as the capacitors  301  and  302 , diodes  303  and  304 , and Zener diode  305 . A charge is temporarily supplied from the capacitor  302  at the moment at which the MOSFETs  201  and  202  are turned on. In this way, as the power needed to drive the MOSFETs  201  and  202  is extremely low on average, unlike a device that carries out high frequency switching like the switching device  7 , it is possible to configure the drive power supply of the MOSFETs  201  and  202  more easily than with a method such as using, for example, an isolated DC/DC converter. 
     The power supply device of the invention can be utilized in applications that supply a constant voltage to various kinds of load such as information and communication instruments, even when there is an instantaneous voltage drop in an alternating current power supply voltage, using an instantaneous voltage drop compensation function.