Abstract:
A control system for an electric motor including an inverter for providing power to the electric motor, a controller for controlling the inverter, a first motor speed control block in the controller injecting a high frequency signal into the electric motor to determine the speed and position of the electric motor, a second motor speed control block in the controller detecting the back electromotive force to determine the speed and position of the electric motor, and a transition control block in said controller to vary operation between the first motor speed control block and the second motor speed control block.

Description:
TECHNICAL FIELD  
         [0001]    The present invention relates to the control of electric motors. More specifically, the present invention relates to a method and apparatus for position sensorless control of an electric motor.  
         BACKGROUND OF THE INVENTION  
         [0002]    Traditional motor control systems normally include a feedback device or position sensor such as a resolver or encoder to provide speed and position information for a motor. Feedback devices and associated interface circuits increase the costs of a motor control system, and these costs may become prohibitive in high volume applications such as automotive applications. Additionally, a position sensor and its associated wiring harness increase the complexity and assembly time of an electric drive system in a vehicle.  
           [0003]    Electric vehicles powered by fuel cells, batteries and hybrid systems that include electric motors are becoming more common in the automotive market. As production volumes for electric vehicles increase, the cost of feedback devices and associated interface circuits will become significant. Automakers are under intense market pressure to cut costs and reduce the number of parts for a vehicle. The removal of a feedback device for an electric motor control system will lead to significant cost reductions for an electric vehicle.  
           [0004]    Hybrid electric and electric vehicles today utilize numerous electric motor control technologies such as the vector control of electric motors. A vector motor control scheme is a computationally intensive motor control scheme that maps the phase voltages/currents of a three-phase motor into a two axis coordinate system. The structure used to excite an electric motor using a vector control scheme is a typical three-phase power source inverter including six power transistors that shape the output voltage to an electric motor. Vector control requires rotor position information, which is normally obtained via a feedback device or position sensor. The objective of the position sensorless control is to obtain the rotor position information utilizing electromagnetic characteristics of an AC machine, eliminating the position sensor and its associated interface circuits.  
         SUMMARY OF THE INVENTION  
         [0005]    The present invention is a method and apparatus for a sensorless control system used in electric and hybrid electric vehicle powertrain applications. The motor control system of the present invention preferably utilizes spatial variation of machine inductance that results from a high frequency signal injection at a relatively low speed (&lt;10% of rated machine speed) and a back electromotive force (EMF) of an AC machine at a relatively high speed (&gt;5% of rated machine speed). While machine speed range limits have been described with reference to a low speed and high speed, it is considered within the scope of the present invention to utilize the spatial variation and back EMF-based methods of the present invention at any speed of an electric motor.  
           [0006]    The present system further includes an initial rotor polarity detection method used to detect the polarity of the rotor magnet during static and dynamic conditions. The present invention will work over the entire torque/speed operating plane, including a standstill and zero stator frequency conditions. The low-speed feedback observer tracks the absolute difference between the D- and Q-axis high frequency impedances to avoid sensitivity to generated harmonics during the stator current transients. At high speed, the present invention uses a full order closed loop speed observer. The rotor speed for the high speed tracking is estimated using a proportional-integral (PI) type controller. The closed form of this observer makes it less sensitive to the parameter variations and allows faster dynamic performance. A transition algorithm provides high level control to supervise the operation of the low- and high-speed sensorless control methods/observers of the present invention. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0007]    [0007]FIG. 1 is a block diagram of a control system in the present invention.  
         [0008]    [0008]FIG. 2 is a block diagram of the low-speed rotor electrical speed/position estimation method of the present invention.  
         [0009]    [0009]FIG. 3 is a block diagram of the initial polarity detection method of the present invention.  
         [0010]    [0010]FIG. 4 is a block diagram of the high speed rotor electrical speed/position estimation method of the present invention.  
         [0011]    [0011]FIG. 5 is a state diagram of the transition method of the present invention. 
     
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT  
       [0012]    [0012]FIG. 1 is a diagrammatic drawing of a preferred embodiment of a control system  10  of the present invention. The control system  10  is illustrated as a sequence of block diagrams that represent software executed in a controller, microprocessor, or similar device to control an electric motor  12 . In the preferred embodiment of the present invention, the controller is a vehicle powertrain controller controlling the electric motor  12 , but any other motor control application is considered within the scope of the present invention. The electric motor may comprise motor technologies such as synchronous reluctance motors, induction motors and interior permanent magnet motors. The input to the control system is a torque command T e  generated by the vehicle controller. The torque command T e  is processed by a torque linearization model  14  to generate a corresponding stator current I s  required to develop the desired electromagnetic torque in the motor  12 . The stator current generated at bock  14  is then passed to an optimum torque per amp block  16 . Block  16  processes the commanded stator current and decomposes it into the respective D and Q axis components of current command (I dse1  and I qse ) to provide the maximum torque for the given stator current amplitude.  
         [0013]    The current command I dse1  is added to a field weakening component I dse     —     fw  generated at summing junction  18  to generate the final D axis current command I dse . The field weakening component I dse     —     fw  is generated by a field weakening block  20  using the measured DC link voltage V dc , commanded output voltages V qss  and V dss , and rotor angular velocity ω r . Summing junction  22  subtracts the feedback current I qse     —     fb  from the Q axis current command I qse  to obtain the error of the Q axis current regulator. Summing junction  24  subtracts the feedback current from I dse     —     fb  from the D axis current command I dse  to obtain the error of the D axis current regulator. The errors generated by the summing junctions  22  and  24  are used by a synchronous current regulator block  26  to control the synchronous frame voltage commands V dse  and V qse .  
         [0014]    Block  28  uses the estimated rotor angular position θ r  to convert the synchronous frame voltage commands V dse  and V qse  to the stationary frame voltage commands V dss1  and V qss1 . The high frequency voltage signals V dss     —     inj  and V qss     —     inj  are added to the stationary reference frame voltage commands by the summing junctions  30  and  32 , resulting in the final voltage commands V dss  and V qss . The voltage source inverter  34  processes the final voltage commands V dss  and V qss  to generate the actual phase voltages applied to the motor  12 . The phase currents are measured and processed by a three-phase to two-phase transformation block  36 . The outputs of the block  36  are stationary frame currents I dss  and I qss . A stationary to rotating frame transformation block  40  uses the stationary frame currents I dss  and I qss  and the estimated rotor angular position θ r  to generate synchronous reference frame feedback currents I dse     —     fb  and I qse     —     fb .  
         [0015]    The present invention includes sensorless control of the rotor speed and position that includes: a low-speed rotor angular position estimation method/observer at block  42 ; an initial rotor polarity detection method at block  43 ; a high speed rotor angular position estimation method/observer at block  44 ; and a transition algorithm at block  46  to seamlessly merge the low- and high-speed estimation methods.  
         [0016]    Block  42  of FIG. 1 represents the low-speed estimation method of the present invention. FIG. 2 shows a detailed block diagram implementation of block  42  to estimate rotor electrical position during low-speed operations as described above. The low-speed estimation method is used to estimate rotor electrical position during zero and low-speed operations (preferably &lt;10% of rated machine speed but any machine speed is considered within the scope of the low-speed estimation method of the present invention). The estimation of the rotor electrical position is performed by injecting a high frequency voltage signal on an estimated D axis of the machine. The fluctuating high frequency signal in a synchronously rotating reference frame with the fundamental stator frequency is used to detect an asymmetry of the spatial impedance in an AC machine. An asymmetry of the spatial impedance is caused by salient construction of the rotor of the machine or induced magnetic saturation in the machine.  
         [0017]    If the high frequency voltage signal is injected on the estimated D axis, the orthogonal component of the current measured at the estimated reference frame can be used as an error signal as shown by equation (1).  
               (           i   dsi                m                 i   qsi                m             )     =       (             y   avg     +       y   diff        cos                 2                   θ   err                 -     y   diff          sin                 2                   θ   err                   -     y   diff          sin                 2                   θ   err               y   avg     -       y   diff        cos                 2                   θ   err               )          (           v   dsi   m               v   qsi   m           )               (   1   )                               
 
         [0018]    where y avg =(z q   e +z d   e )/2z d   e z q   e  and y diff =(z q   e −z d   e )/2z d   e z q   e . If the voltage signal is injected on the estimated D-axis (v dst   m =V inj  sin ω t  and v qst   m =0), then in the Q-axis current signal the diagonal component disappears and the off-diagonal component appears as shown in equation (2). If resistive components are much smaller than inductive components (r d   e , r q   e &lt;&lt;x d   e , x q   e ) at the high frequency and also the impedance difference of the reactive component is much larger than that of the resistive component (|x d   e −x q   e |&gt;&gt;|r d   e −r q   e |), then equation (2) can be simplified as shown in equation (3) in quasi-steady-state.  
               i   qsi                m       =           (       -     y   diff          sin                 2                   θ   err       )     ·     V     i                 n                 j            sin                   ω   h        t     =         [         (         r                d   e     -     r   q   e       )     +     j        (       x   d                e       -     x   q                e         )             (       r   d                e       +     j                   x   d                e           )     ·     (       r   q                e       +     j                   x   q                e           )         ]     ·     
          (       V     i                 n                 j          sin                 2        θ   err       )     ·   sin                     ω   h        t               (   2   )                   i   qsi                m       ≈       -   j                x   d                e       -     x   q                e               x                d   e          x   q                e           ·     (       V     i                 n                 j          sin                 2        θ   err       )     ·   sin                     ω   h        t       =         -         x   d   e     -     x   q   e           x   d   e          x   q   e           ·                
          (       V     i                 n                 j          sin                 2        θ   err       )     ·   cos                     ω   h        t             (   3   )                               
 
         [0019]    Multiplying the orthogonal signal with respect to the injected signal results in the DC quantity of the error signal for the tracking controller. After low-pass filtering, the DC quantity can be obtained as shown in equation (4).  
                   ɛ   =       LPF        [       i   qsi                m       ×     (       -   cos                     ω   h        t     )       ]       =         -       Y   2     2          sin                 2        θ   err       ≈       -     Y   2            θ   err                           where                   Y   2       =       -         x   d                e       -     x   q                e             x   d                e            x   q                e             ·                                (   4   )                               
 
         [0020]    Referring to FIG. 2, block  50  converts the stationary frame currents I qss  and I dss  to the estimated synchronous reference frame current I qsm . Block  52  comprises a second order band pass filter to allow only the injection high frequency signal (preferably in the range 300 to 1000 Hz) to be processed at multiplying junction  54 . Junction  54  multiplies the output of the BPF of block  52  by the term −cos(ω inj t) to extract the DC component of the error signal. Block  56  comprises a second order low pass filter to remove high frequency harmonics from the signal and output the term ε. ε is an error signal defined in equation (4).  
         [0021]    Block  58  is a third order position observer that processes the error term ε. ε is processed by proportional control block  60 , integral control block  62 , and feed-forward control block  64  to generate outputs. The integral and proportional outputs of blocks  60  and  62  are summed at summing junction  66  and processed by block  68  to generate and estimate speed ω r     —     low . The output of the feed-forward gain block  64  is processed by a limiter block  70  and then fed forward to summing junction  72  to be added to the speed output of block  68 . Block  74  processes the output of summing junction  72  to generate the term θ r     —     low  which is the estimated angular position of the rotor at low speed.  
         [0022]    [0022]FIG. 3 is a detailed block diagram implementation of the block  43  used to detect initial rotor magnet polarity. The stationary to rotating reference frame block  80  converts the stationary frame currents I dss  and I qss  to the synchronous reference frame currents I dse  and I qse  using θr. Only the D axis current I dse  is used in the initial rotor polarity detection method. I dse  is passed through a band-pass filter  82  which filters out all but the second harmonic of the injection frequency of the I dse  current. The output of the band-pass filter  82  is I dse     —     bp . The signal I dse     —     bp  is demodulated by multiplying it with the term sin(2ω inj t−φ) using the multiplier block  84 . The resultant signal I d1  will contain a DC component and a high frequency component. The low-pass filter block  86  filters out the high frequency component of I d1 , leaving only the DC portion I d . The signal I d  contains the information on the polarity of the rotor magnet with respect to the estimated machine D axis. Condition block  88  determines the polarity of the estimated position using the sign of the signal I d . This condition may be evaluated only once during the start-up sequence. If the sign of I d  is negative, 180 degrees is added to the estimated rotor position.  
         [0023]    [0023]FIG. 4 is a detailed block diagram implementation of the high speed estimation method of block  44  seen in FIG. 1. Block  89  is an estimator for the D and Q axis BEMF voltages E dq  using measured synchronous frame currents I dqse  and commanded synchronous frame voltages V dqse . Commanded voltages V dqse  are multiplied by block  90 , which represents the matrix “B” in the following Equation 5. Blocks  94 ,  98 , and  108  represent the matrix “A” in Equation 5. The output of summing junction  92  is integrated by block  96  resulting in an estimated synchronous reference frame currents I dqse     —     hat . The output of integrator block  96  is supplied to block  94 , junction  100 , and block  102 . Multiplier junction  100  multiplies the output of block  96  by the estimated angular rotor speed ω r  and supplies its output to block  98 . Summing junction  102  compares measured and estimated currents to generate an error signal, which is in turn supplied to block  104 . Block  104  is a gain matrix representing matrix “G” in Equation 5. Block  106  integrates the output of  104  to generate estimated D and Q axis BEMF voltages. Summing junction  92  adds the outputs of blocks  90 ,  94 ,  98 ,  104 , and  108  to complete state matrix “x” in Equation 5.  
         [0024]    A closed loop full state observer of block  89  can be expressed by the following equation:  
           {circumflex over (x)}=A{circumflex over (x)}+Bu+G ( y−C{circumflex over (x)} )  (5)              where                   x   ^       =       [             i   ^     dse             i   ^     qse             E   ^     d             E   ^     q           ]     T       ,     y   =       [           i   dse           i   qse           ]     T       ,     
          A   =     [           -       r   s       L   d                 ω   r            L   q       L   d               1     L   d           0               -     ω   r              L   d       L   q               -       r   s       L   q             0         1     L   q               0       0       0       0           0       0       0       0         ]       ,     B   =     [           1     L   d           0           0         1     L   q               0       0           0       0         ]       ,     
          C   =     [         1       0       0       0           0       1       0       0         ]       ,     G   =     [           g   11           g   12               g   21           g   22               g   31           g   32               g   41           g   42           ]                                            
         [0025]    Estimated back EMF E dq  is used to generate the speed and rotor electrical position using blocks  110 - 118 . Block  110  is used to generate the proper scaling and polarity of the error signal to the PI block  112 . If the estimate is correct, E d  is equal to zero. However, if E d  is non-zero, then it can be used as the error signal to the PI block  112 , resulting in an estimated rotor angular velocity ω r  Integrator block  114  generates the estimated rotor position θ r  based on the estimated rotor angular velocity ω r . Rotor electrical position correction controller  118  is used to compensate any error in the estimate due to the non-linearity of the system through summing junction  116 .  
         [0026]    [0026]FIG. 5 illustrates a state flow diagram for the transition method block  46  of the present invention that provides for a smooth transition between high- and low-speed rotor angular position and speed estimation methods. The transition algorithm described in FIG. 5 provides high level control to supervise the operation of the low- and high-speed sensorless control methods. Upon power-up of the controller, the algorithm begins with the start module  120 , which performs general initialization functions. In the case of a permanent magnet machine, block  122  is used to determine the initial polarity of the rotor magnet (i.e., north/south orientation). Once the initial rotor polarity detection is complete, the algorithm enters a low-speed mode  124 , and remains there until the conditions described in condition block  126  have been satisfied. When the condition block  126  becomes true, the control is passed to the high-speed mode  132 . Control remains in the high-speed mode until the conditions described in condition block  130  have been satisfied. When the condition block  130  becomes true, the control is returned to the low-speed mode  124 . The threshold speeds ω LH  and ω HL  are chosen with sufficient separation to prevent multiple transitioning back and forth between modes. The injection voltage magnitude at block  128  is programmed as a function of rotor speed. At low-speed, the injection voltage is held constant. If speed exceeds a predefined threshold, the injection voltage is reduced linearly with respect to speed. The injection voltage is clamped to zero during high-speed mode.  
         [0027]    It is to be understood that the invention is not limited to the exact construction illustrated and described above, but that various changes and modifications may be made without departing from the spirit and scope of the invention as defined in the following claims.