Abstract:
A communication system comprises a twisted pair communication link operably coupled to at least two driver stages for providing at least two independent input signals on the twisted pair communication link. The at least two independent input signals on the twisted pair communication link are summed and input to a comparator arranged to compare the summed signal to a reference value. The output of the comparator is input to the at least two driver stages. The outputs from the at least two driver stages are summed and fed back and summed with one or more of the independent input signals. 
     In this manner, adverse effects due to non-ideal symmetry between components in a twisted pair communication link, such as a Controller Area Network system, are reduced.

Description:
FIELD OF THE INVENTION 
     The preferred embodiment of the present invention relates to reduction of emission level in a twisted pair communication system. The invention is applicable to, but not limited to, provision of a high speed two-wire CAN network. 
     BACKGROUND OF THE INVENTION 
     In the field of vehicle electronics, the use of a high speed controller area network (CAN) is becoming ever more prevalent. In a typical automotive application, the CAN provides a two-wire multiplex communication link that can be routed around the vehicle. Thus, the CAN provides a simple mechanism for a vehicle processing unit to communicate effectively with remote electrical/signal processing units, e.g. vehicle light modules, braking system, airbag modules, etc. 
     The CAN specifications for road vehicles are defined by the International Standards Organisation (ISO), as described below.
         (i) ISO 11898-2: high speed physical layer—part 2;   (ii) ISO 11898-3: low speed fault tolerant physical layer—part 3;   (iii) ISO 11898-4: time trigger CAN; and   (iv) ISO 11898-5: high speed physical layer with low power mode and wake up.       

     The CAN specifications support communication over, say, a ten metre length. However, in requiring the CAN bus to support communication over this length, the long wires act as an antenna and as such are effectively subject to automotive electrical transients, as well as industrial transients, such as electro-magnetic interference (EMI) and electro-static discharge (ESD). 
     Furthermore, as the wires can be very long, the control of the slew rate of signals routed by the wires is also known to be very critical to avoid any EMC emission. In addition, in order to operate the CAN bus successfully, in a problematic vehicle environment; a CAN transceiver must also be able to withstand high voltage transients. 
     Consequently, it is important to guarantee a good control and matching of the respective devices between the switching of the two CAN wires. To guarantee good matching of the slew rate between the respective wires, the two wires (i.e. the high-side CAN (CAN-H) and the low-side CAN (CAN-L)) must be matched in performance and controlled equally. 
     A typical CAN driver circuit is illustrated in  FIG. 1 . The CAN driver circuit comprises a digital transmit input signal  102  that is input to both the CAN-H driver  104  and a CAN-L driver  106 . In order to achieve both high-speed and symmetry of operation, low-voltage matched components are generally used. The CAN-H driver utilizes, say, a pnp transistor  130  as an active device operably coupled to Vcc  108 , where as the CAN-L driver utilizes, say, a npn transistor  132  as an active device, operably coupled to ground  110 . 
     The respective outputs  118 ,  120  from the CAN-H and CAN-L drivers  104 ,  106  are input  114 ,  116  to a comparator  112 . The output from the comparator  112  is a ‘receive’ digital output signal  122 . Thus, as will be appreciated by a skilled artisan, the driver circuits that control the signals on the CAN-H wire  118  and CAN-L wire  120  need to be carefully matched, to ensure that the CAN-H driver  104  and CAN-L driver  106  are adjusted to switch between the CAN-H and CAN-L wires  118 ,  120  in phase. A key aspect to using such a twisted cable is to ensure the current contained in the wires (in both directions) is of equal amplitude and of opposite sign. In this manner, the magnetic field produced, which is proportional to the current, is substantially zero. Thus, electro-magnetic interference due to the current in the twisted pair of cables is minimized. Clearly, any asymmetry between the current in the two wires produces a magnetic field, which is highly undesirable. 
     In addition, it is known that the common-mode CAN-H bus wire and CAN-L bus wire must also be constant during the switching transition, i.e. when a signal appears on the CAN-H and CAN-L wires, and when it is taken off. Thus, the ΔI needs to be minimized during the transitions otherwise electromagnetic interference is created. In effect, there are two types of common-mode configuration:
         (i) Current: where the current value of both the CAN-H and CAN-L wires should be equal but of opposite sign; and   (ii) Voltage: which should be equal, given that the for high speed operation the bus impedance is specified as 60 ohms in the CAN standard.       

     Furthermore, to avoid EMC emission, the slew rate applied to signals on the CAN-H bus wire  118  and the CAN-L bus wire  120  must be controlled and matched. The slew rate is a function of the temperature (delay) and of the load. Consequently, the slew rate is difficult to control accurately. 
     A graphical example of how difficult it is to achieve a good match between the high-side driver and the low-side driver is shown in  FIG. 2 . Here, the transmit waveform signal  202  is shown with a slight offset to the receive waveform signal  222 . If the respective driver circuits are symmetrical, the CAN-H and CAN-L signals in waveform  218  are also symmetrical, resulting in the summation of CAN-H (nominally 3.5V) and CAN-L (nominally 1.5V) values to be flat, at approximately 5V. The summation of the CAN-H and CAN-L signals is often referred to as the ‘common mode’. However, when the driver circuits are not completely matched, the symmetry between CAN-H and CAN-L during transitions is not met. This lack of matching results in a so-called common-mode glitch  230  (say a variation of the order of 120 mV), which is noticeable upon the summation of the CAN-H and CAN-L values. 
     A solution to this problem is illustrated in the known prior art circuit  300  of  FIG. 3 , with the use of multiple drivers  316 ,  318 ,  320 ,  322  for the CAN-H and drivers  324 ,  326 ,  328  and  330  for CAN-L. A series of very fast switches  336 ,  338 ,  340 ,  342 ,  344 ,  346 ,  348  and  350  are operably coupled to respective serial resistances, where each driver operation is controlled by a fixed delay  304 ,  306 ,  308 ,  310 ,  312  and  314 . 
     In this regard, the slew rate is fixed by the delay elements  304 ,  306 ,  308 ,  310 ,  312  and  314 , the series resistances and the load capacitance. Accurate selection of these components ensures a good match. 
     However, with a high voltage range on the output, non symmetrical clamping is (due to the inherent nature of the components) introduced in series with resistance. These high voltage components can be designed to be somewhat symmetrical at low frequencies. However, they will exhibit asymmetry at higher frequencies, say above 100 KHz. Thus, asymmetry of signals between the two wires will generate common-mode glitches at a frequency of above 100 KHz. In effect, a new common-mode (i.e. the summation of the values of the CAN-H and CAN-L wires) exists at each frequency of operation. 
     European Patent EP 0955750, titled “Line driver with parallel driver stages” by Texas Instruments, as well as European Patent EP0763917 titled “Line driver with pulse shaper” by Lucent Technologies Inc. U.S. Pat. No. 5,194,761 illustrates prior art CAN arrangements. 
     However, it is recognised that, in order to sustain higher and higher voltage levels during fault conditions, it is no longer possible to use low voltage components. Thus, instead, high voltage components are required to be used, particularly components that exhibit higher parasitic effects. Notably, and problematically, the parasitic components exhibit different characteristics for the high side driver (CAN-H) and the low side driver (CAN-L) outputs. Consequently, the common-mode performance becomes degraded with those high voltage components. 
     Thus, a need exists for an improved twisted-pair based communication system, apparatus and method therefor, particularly to drive CAN-H and CAN-L bus wires. 
     STATEMENT OF INVENTION 
     In accordance with preferred aspects of the present invention, there is provided a communication system, an apparatus and method therefor to reduce emission levels due to non-ideal symmetry between components in twisted pair paths, as defined in the appended Claims. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates a known CAN driver circuit; 
         FIG. 2  shows a graphical example illustrating the difficulty in achieving a good match between a CAN-H driver and a CAN-L driver is a CAN bus system; and 
         FIG. 3  illustrates a known multiple CAN driver circuit that aims to provide a good match between a CAN-H driver and a CAN-L driver. 
       Exemplary embodiments of the present invention will now be described, by way of example only, with reference to the accompanying drawings, in which: 
         FIG. 4  illustrates a CAN driver circuit in accordance with the preferred embodiment of the present invention; and 
         FIG. 5  illustrates graphically how the circuit of  FIG. 4  provides a reduction in the common-mode error in accordance with the preferred embodiment of the present invention. 
     
    
    
     DESCRIPTION OF PREFERRED EMBODIMENTS 
     Referring now to  FIG. 4 , a CAN driver circuit is illustrated in accordance with the preferred embodiment of the present invention. Notably, the CAN-H signal  406  and the CAN-L signal  402  are input to a summation function  408 . It is envisaged that the summation function may be implemented in a variety of ways, such as a summing block, discrete logic gates, AC coupling capacitors, etc. The CAN-H signal  406  and the CAN-L signal  402  are thus summed and input  414  to a fast comparator  418  at, say, a negative input port  414 . 
     A reference voltage level  412  is applied to the positive input port  416  of the comparator. The comparator  418  compares the summation of CAN-H and CAN-L signals with a reference voltage  412 , which in the preferred embodiment is set at 5V. The comparator  418  outputs the comparison to both a CAN-H driver stage  424  and a CAN-L driver stage pair  422 ,  424 . The CAN-H driver stage  424  is coupled to a supply voltage  426  and the CAN-L driver stage  422  is coupled to ground  428 . 
     Although the preferred embodiment of the present invention is described with respect to using NMOS and PMOS transistors  422 ,  424  at the output of the comparator  418  to generate a CAN-L feedback signal, it is envisaged that these devices may be replaced by equivalent functions, such as switches or current sources or current sources with resistors coupled thereto, etc. 
     Notably, the output from both the CAN-H driver stage  424  and the CAN-L driver stage  422  is summed and provided as a feedback signal to either the CAN-H or CAN-L (or indeed both) signal(s). In comparing the summation of ‘CAN-H+CAN-L’ to a reference voltage applied to the comparator, the output of the comparator flips as soon as the summation goes above or below the reference value. Thus, dependent upon the comparison of the common-mode (summation of CAN-H and CAN-L, nominally 5V) value to the reference value (of 5V), current is either drawn from the CAN-L or pushed into the CAN-L feedback path to compensate for the difference. In this manner, the value of ‘CAN-H+CAN-L’ is either decreased or increased via the feedback path. 
     The speed of the comparator defines the reaction time of this loop, and thus the difference between the summation of ‘CAN-H+CAN-L’ and the reference voltage. 
     Thus, in this manner, a mechanism to improve the performance of the CAN circuit  400  is described by summing the CAN-H signal  406  and CAN-L signal  402  and using a fast, continuously-operating comparator  418 , e.g. one that provides sufficient speed and low offset between the two wires. Thereafter, the output from the comparator  418  is fed back to one or both of the CAN signals  406 ,  402 . A typical (CMOS) comparator error of 10 mV will result in a common-mode glitch of 10 mV. 
     Furthermore, in this implementation, the current compensation is limited to, say, 10% of the total current, to avoid an adverse affect on the performance (e.g. slew rate, propagation time, etc.) when the compensation system is disturbed by EMC perturbation (BCI, DPI, etc.). 
     In a CAN system, the transition rise time and fall time of the CAN-H and CAN-L signals has to be typically between 15 nsec and 35 nsec at a speed of 1M baud. Thus, it is envisaged that if the two CAN wires cannot be accurately matched using a single driver stage, a series of respective driver stages may be used, where the series of small stages are located in parallel, as per the known system in  FIG. 3 . 
     In this regard, several stages may be sequentially triggered in parallel. For example, if 10 stages are located in series, for each of the parallel CAN-H and CAN-L paths, the comparator may need to switch between the stages every 1.5 nsec to 3.5 nsec per transition. Hence, in this regard, a switching performance of the comparator of between 1 nsec and 5 nsec is likely to be acceptable. 
     Although, the preferred embodiment of the present invention is described with reference to a continuously-operating comparator, it is envisaged that the inventive concept is equally applicable to a ‘time-discrete’ comparator, as would be understood by a skilled artisan. In this context, a ‘time-discrete’ comparator would encompass a ‘sample &amp; hold’ type of comparator, whereby values of the CAN-H and CAN-L signals are sampled and held and then compared in a time-discrete manner. Advantageously, a number of ‘sample &amp; hold’ comparators offer an offset-cancellation technique. The use of an offset-cancellation technique allows the removal of any common-mode glitch that occurs due to the comparator offset. 
     It is also envisaged that known cancellation techniques may be applied to continuously-operating comparators such that they also can implement cancellations of common-mode glitches due to a comparator offset. 
     In an enhanced embodiment of the present invention, it is envisaged that the common-mode glitch problem is only applied during transition stages of the CAN system. In this regard, it is envisaged that the feedback circuitry is enabled at the start of a transition and disables after the transition has ended. Advantageously, employing such a time-multiplexed arrangement assists in minimising disturbances in the CAN system caused by the aforementioned circuitry. 
     The use of a fast comparator in this manner compensates for any real-time imbalance between the signal values on the CAN-H and CAN-L wires, thereby facilitating a dramatic reduction in emission level by limiting the common-mode glitch that occurs during transition on the bus lines. 
     Advantageously, the current output capability of the CAN compensation circuit is limited to a certain (low) value, in contrast to the main CAN-L and CAN-H capabilities. Indeed the slew rate on (CAN-H or CAN-L) is directly proportional to the output current. Thus, by using a series of smaller MOSFETs, any variation of current in, say, the CAN-L line causes minimal modification to the slew rate. Consequently, it is an advantage to limit the current transition, particularly in vehicular applications, to minimise the effect of radiated emissions due to, say, mobile antenna(s) or portable TVs. 
     At each stage transition, the comparator  418  thus compensates the intrinsic error of common-mode ‘CAN-H+CAN-L’ by continuously adding or subtracting current  430  in order to keep a common-mode value substantially equal to the reference value  412 . By ensuring that the summation value of ‘CAN-H+CAN-L’ is made substantially equal to a predefined reference value, any electromagnetic emission resulting from using a twisted pair of wires (as per the CAN system) is effectively reduced, i.e. the magnetic fields generated by each wire are opposed in sign and therefore the resultant value of the summation is substantially ‘zero’. 
     Referring now to  FIG. 5 , a graph illustrates how the circuit of  FIG. 4  provides a reduction in the common-mode error in accordance with the preferred embodiment of the present invention.  FIG. 5  illustrates a transition between CAN-H and CAN-L signals, and in particular the correction current that maintains a low glitch level. 
     Although the preferred embodiment of the present invention has been described with reference to a CAN circuit, it is envisaged that, for alternative applications, the inventive concept may be applied to any communication system that employs twisted pair cabling, such as those used by Ethernet transceivers. 
     It will be understood that the communication system, apparatus and method for reducing emission levels in a two-wire communication system, such as a CAN system as described above, aims to provide at least one or more of the following advantages:
         (i) Improve the non-ideal symmetry on CAN-H, CAN-L drivers;   (ii) Reduce the reliance on matched components within the device used between the two wires;   (iii) The common-mode glitch problem associated with bus transitions is better controlled;   (iv) The preferred embodiment is very robust to process variations, due to the introduction of the feedback loop;   (v) The EMC emission is dramatically reduced; and   (vi) The circuit of the preferred embodiment is such that it is very easy to reuse on other applications/technologies; i.e. substantially no software/hardware modifications are required.       

     In particular, it is envisaged that the aforementioned inventive concept can be applied by a semiconductor manufacturer to any device or integrated circuit for use in any communication system employing twisted cabling. It is further envisaged that, for example, a semiconductor manufacturer may employ the inventive concept in a design of a stand-alone device, such as an Ethernet transceiver, or an embedded module in an application-specific integrated circuit (ASIC) and/or any other sub-system element. 
     Whilst the specific and preferred implementations of the embodiments of the present invention are described above, it is clear that one skilled in the art could readily apply variations and modifications of such an inventive concept. 
     Thus, an improved twisted pair communication system, apparatus and method therefor; to reduce emission levels due to non-ideal symmetry between components in the respective twisted pair paths have been described, wherein the aforementioned disadvantages with prior art arrangements have been substantially alleviated.