Abstract:
A method for providing additional dc inputs or outputs ( 49, 59 ) from a dc-to-ac inverter ( 10 ) for controlling motor loads ( 60 ) comprises deriving zero-sequence components (V ao , V bo , and V co ) from the inverter ( 10 ) through additional circuit branches with power switching devices ( 23, 44, 46 ), transforming the voltage between a high voltage and a low voltage using a transformer or motor ( 42, 50 ), converting the low voltage between ac and dc using a rectifier ( 41, 51 ) or an H-bridge ( 61 ), and providing at least one low voltage dc input or output ( 49, 59 ). The transformation of the ac voltage may be either single phase or three phase. Where less than a 100% duty cycle is acceptable, a two-phase modulation of the switching signals controlling the inverter ( 10 ) reduces switching losses in the inverter ( 10 ). A plurality of circuits for carrying out the invention are also disclosed.

Description:
STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH 
     This invention was made with assistance under Contract No. DE-AC05-00OR22725 with the U.S. Department of Energy. The Government has certain rights in this invention. 
    
    
     BACKGROUND OF THE INVENTION 
     The use of dc-to-ac PWM inverters in adjustable speed drives for controlling loads is becoming increasingly popular. A motor can be driven in an ac synchronous mode or brushless dc (BLDC) mode using a dc link inverter to control three-phase switching of current in the windings of a 3-phase motor. 
     One object of the invention is to increase efficiency and to lower the cost of manufacture by integrating electromagnetic components. 
     When the power supply is a dc source, it is desirable to have other auxiliary dc and ac power components of different voltages operated from the sole dc power source. It may also be desired to convert energy from a low voltage dc supply back to the high power dc power source. There are many dc-to-dc converters that can be used to supplement a main ac inverter, but these are separate components. The present invention is designed for application in an apparatus which combines one or more dc-to-dc or dc-to-ac conversions with a motor being controlled by the main inverter. 
     The increasingly sophisticated application of the contemporary technology often calls for several electromagnetic devices to be used simultaneously. For example, in certain uses of electric motors it is desirable to add various associated devices such as the auxiliary power systems, filters, transformers, and chokes. This creates an opportunity for cost reduction by forming multiple devices from individual components. 
     SUMMARY OF THE INVENTION 
     The present invention provides a method and an electronic motor control that utilize zero-sequence components to provide dc-to-dc power conversion for low voltage dc devices operated in a circuit with a larger component, such as a motor. Such low voltage dc devices can include, but are not limited to, dc-to-dc converters, transformers, filter chokes, ac output power supplies, and smoothing filters for the main stator windings in PWM applications. These devices can be physically integrated, at least in part, with the control, inverter, and motor. The present invention provides the necessary power supply voltages from a single motor control. 
     The zero-sequence voltage or current produced by a PWM inverter can be used to drive a three-phase transformer with the individual phase secondaries of the transformer connected in series to produce a suitable waveform. 
     The invention provides a method for reverse power conversion in which energy is converted from the low voltage dc side back to the high voltage dc power source. 
     One advantage of the invention is that the use of zero-sequence currents does not generate any additional torque. 
     The invention also enables a reduction in the size and reduction in manufacturing cost of the motor drives and accessory power systems. 
     The invention may be applied to induction machines as well to as permanent-magnet (PM) machines and synchronous machines. As used herein, the term “machines” shall include both motors and generators. 
     The invention may be practiced with 2-phase modulation which lowers the switching losses of the main inverter. 
     Other objects and advantages of the invention, besides those discussed above, will be apparent to those of ordinary skill in the art from the description of the preferred embodiments which follows. In the description reference is made to the accompanying drawings, which form a part hereof, and which illustrate examples of the invention. Such examples, however are not exhaustive of the various embodiments of the invention, and therefore reference is made to the claims which follow the description for determining the scope of the invention. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is an electrical schematic diagram of a PWM inverter of the prior art; 
     FIG. 2 is a detail electrical schematic with waveforms showing the development of PWM signals according to the prior art; 
     FIG. 3 is a comparison graph of the signals of FIG.  2  and of the zero-sequence components and of line voltage as a function of time; 
     FIG. 4 is an electrical schematic of the inverter of FIG. 1 with the addition of a zero-sequence circuit; 
     FIG. 5 is an electrical schematic of a modified form of the circuit of FIG. 4; 
     FIGS. 6 a - 6   c  are graphs of sine wave modulating signals, of the zero-sequence components, and of a non-zero line-to-line voltage, respectively, as a function of time; 
     FIG. 7 is a graph of sine wave modulating signals vs. time for a 2-phase modulation; 
     FIGS. 8 a - 8   c  are graphs of the of sine wave modulating signals, the zero-sequence components, and the line-to-line voltage for a 2-phase modulation; 
     FIG. 9 is an electrical schematic diagram with the addition of a low voltage dc supply connected to receive zero-sequence components from an inverter; 
     FIG. 10 is the electrical schematic diagram of FIG. 9 with only one power switch in the fourth leg of the inverter circuit and with capacitors added in each leg of the low voltage dc supply; 
     FIG. 11 is a test plot of the dc current supply waveforms developed with a circuit according to FIG. 10 with a 2-phase modulation; 
     FIG. 12 is a test plot of the dc current supply waveforms developed with a circuit according to FIG. 10 with a 3-phase modulation; 
     FIG. 13 is an electrical schematic diagram of an inverter according to FIG. 9 with multiple dc supply outputs added; 
     FIG. 14 is an electrical schematic diagram of an inverter according to FIG. 10 with multiple dc supply outputs added; 
     FIG. 15 is a simplified version of the electrical schematic diagram of FIG. 13; 
     FIG. 16 is a simplified version of the electrical schematic diagram of FIG. 14; and 
     FIG. 17 is an electrical schematic diagram of an inverter with the addition of a reverse charging circuit. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     FIG. 1 shows a conventional 3-phase inverter  10  that is fed from a dc voltage source  11 ,  12  with a line-to-line voltage, V dc . The middle potential of V dc  is labeled as o. The 3-phase wye-connected circuit of impedances, Z, is electrically connected to the inverter legs at points a, b, and c, respectively. The gate signals of the six power electronic switching devices  13 ,  14 ,  15 ,  16 ,  17  and  18  control the potentials of the a, b, and c points individually. Each switching device  13 - 18  has a diode  19  connected in parallel to protect it from reverse voltages and to provide a discharge path when the switch is to be turned off. The gate signals are provided under programmed control of a microelectronic processor (not shown) as is well known in the art. If the switching devices  13 - 18  are insulated gate bipolar transistor (IGBT) power electronic switching devices, these can be turned on with a few volts of a high gate signal, and turned off with a low gate signal. Consequently, the a, b, and c points can be connected to either the positive (+) side of the dc bus or to the negative (−) side of the bus. 
     It is known that the PWM gate signals of the power electronic switching devices can be obtained according to the principle of signal modulation through a triangular carrier wave, C, as shown in FIG. 2. A sine-wave modulating signal, M, is compared with the triangular carrier wave, C. It results in a multiple-pulse gate-signal waveform PWM for turning on or off of the two power electronic switches of one leg of the inverter  10 . If the uppermost power electronic switch in a leg is “on,” the bottom one must be “off,” or vice versa, to avoid a short in one branch of the circuit. 
     FIG. 3 shows that when the phase modulating signal is zero, the zero-sequence phase voltages, V ao , V bo , and V co , are not zero. They have the potentials of either the positive (+) or the negative (−) potential of the dc bus. The line voltage, V ab  does not contain any zero-sequence component produced by the carrier signal, thus the line voltage such as, V ab , is zero. As seen in FIG. 3, the three zero-sequence phase voltages, V ao , V bo , and V co  are in phase. They are the zero-sequence components produced by zero-sequence switchings between positive and negative values. 
     In FIG. 4 the currents of these zero-sequence components, V ao , V bo , and V co , can be collected from the neutral, N, of a 3-phase network. Thus, the zero-sequence circuit is partially formed by the switches  13 - 18 , the 3-phase network connected in a wye configuration, and an impedance Zo. The neutral current going through an impedance, Zo, can be controlled through a fourth leg with power switches  22 ,  23  added to the main three-phase inverter  10 . For each phase, the power switching devices  13 - 18  of the inverter  10  also act with the fourth leg to form an H-converter to control the zero-sequence current. 
     FIG. 5 shows that the circuit given in FIG. 4 can be simplified. Only one additional power electronic switching device  23  is necessary in the fourth leg for controlling the zero-sequence current, provided that a diode  24  is used in place of switch  22  and provided that a capacitor, C 1 , is also added to each phase of the 3-phase network. 
     It should be mentioned that the zero-sequence components in the three phases shown in FIG. 3 are identical. The zero-sequence components associated with V ao , V bo , and V co  would be affected, in the event that the modulating signals were not zero. 
     FIGS. 6 a - 6   c  show when the modulating signals M are not zero, and when there are three sine waves, the switchings of the V ao , V bo , and V co , are not identical. For example, V ao  contains a sine wave component M and pulses at the frequency of the carrier wave C. The value of V ab  is no longer zero. The zero-sequence components V ao , V bo , and V co  are modulated by the sine waves M. 
     FIG. 7 illustrates that the sine wave modulating signals, V sina , V sinb , and V sinc , can be further modulated to produce two flattened regions of magnitude per cycle by adding a V add  signal as those shown in FIG.  7 . This is referred to as a two-phase modulation, which by itself, is known in the art. 
     FIG. 8 a  shows that the two-phase modulating waves M are more non-sinusoidal than the sinusoidal, distributed modulating signals M shown in FIG.  6 . The two non-switching regions per cycle represented by the flat portions of the waves M shown in FIG. 8 a  reduce the duty cycle per phase to two-thirds duty-cycle waveforms. These duty cycles produce asymmetrical zero-sequence switching components V ao , V bo , and V co , in FIG. 8 b  and line-to-line voltage V ab  seen in FIG. 8 c . However, the sum of the three phases can make the envelope of the zero-sequence-switching output waves more even as compared with those of the single-phase outputs produced by the circuits shown in FIGS. 15 and 16. 
     In order to obtain auxiliary dc output voltages different from a dc power source, three basic functions must be provided. They are: 1) a switching function that changes dc to ac, 2) a transformer function that steps down or steps up the ac voltage, and 3) a rectifier function that converts ac to dc. Additional ac outputs can also be obtained from the first two functions. 
     FIG. 9 shows an example of a motor control circuit having an inverter  10  and a dc-to-dc low voltage output power source  41 ,  42 . The power switching devices  13 ,  14 ,  15 ,  16 ,  17 ,  18  of the three legs of the main inverter  10  produce adjustable currents in the motor main windings (not shown) as well as the controllable zero-sequence-switching currents in the three phase transformers  42   a ,  42   b ,  42   c  for the auxiliary converter  42  for a dc low voltage power output at output terminals  49 . The three-phase transformers  42   a ,  42   b ,  42   c  of the auxiliary converter  42  can be built into a motor or be separated from the motor. The main windings of the motor can also be used as the primary windings of the transformers. This is based on the known fact that the zero-sequence electric components are not coupled with the positive-sequence electric components that drive the motor. In FIG. 9, two switching devices  43 ,  44  are used in the branch of the circuit added to the inverter  10 . As seen in FIG. 10, only one power-switching device  44  (such as an IGBT) for each auxiliary voltage output  49  is required in the fourth branch of the circuit, along with diode  47 , provided that a capacitor C 2  is added to each of the transformer circuits  42   a ,  42   b  and  42   c . These circuits utilize the zero-sequence switching of the switching devices  13 - 18  of the main inverter  10 . The transformers  42   a ,  42   b ,  42   c  provide for stepping up, or in this case, stepping down of the ac voltage of the zero-sequence switching currents. These voltages are then rectified by the rectifier  41  to provide a dc voltage at output terminals  49 . 
     FIG. 11 shows the dc charging current waveforms of the system shown in FIG. 10 at 100, 55, and 0 amperes, respectively, under a full 2-phase modulation. 
     FIG. 12 shows the dc charging current waveforms of the circuit shown in FIG. 10 at 102 and 0 amperes, respectively, under a conventional 3-phase modulation. The 2-phase modulation in FIG. 11 lowers the switching losses of the main inverter  10  in the circuits described above, but may provide less than a 100% duty cycle. 
     As shown in FIG. 13, multiple low-voltage DC outputs  49 ,  59  can also be provided. In FIG. 13, two low-voltage outputs and two low-voltage control branches with pairs of power electronic switches  43 ,  44  and  45 ,  46  are connected in parallel with the inverter  10 . The center output points of the legs are connected to a pair of auxiliary dc converters in one device  50 . This device  50  provides two sets of transformer or motor windings  42   a - 42   c ,  52   a - 52   c , which connect to rectifiers  41 ,  51 . 
     FIG. 14 shows the simplification of FIG. 13 in which the power switches  43 ,  45  can be replaced by diodes  47 ,  48 , respectively, provided that capacitors C 2  are added to the transformer circuits  42   a ,  42   b ,  42   c ,  52   a ,  52   b  and  52   c.    
     If a less than 100% duty-cycle low DC voltage output is acceptable to the user, the circuits of FIGS. 11 and 12 can be simplified to those shown in FIGS. 15 and 16. In FIGS. 15 and 16, the inverter  10  is shown connected to a motor or other loads  60 . Two branches are added to the inverter, in FIG. 15, including pairs of power switching devices  43 ,  44  and  45 ,  46 . A connection is made between the pairs of devices to receive the zero-sequence switchings of the main inverter. These are conducted through a single transformer coil  42 ,  52 , respectively, to provide a single-phase transformer function. The output of the transformers is conducted to the rectifiers  41 ,  51  seen in FIGS. 13 and 14 to complete a single-phase low voltage dc supply output. In this circuit the zero-sequence switchings exhibit a duty cycle that occurs when the magnitude of the fundamental voltage of the main inverter goes up. 
     FIG. 16 is a simplified version of the single-phase conversion of FIG. 15, in which diodes  47 ,  48  are substituted for power switching devices  43 ,  45  in FIG.  15 . 
     FIG. 17 shows an example of a circuit for feeding back a low voltage at outputs  49  to the main power supply  80 . This might allow the conversion of the low voltage power for boosting the main power supply  80  during an emergency situation when the main power supply is sagging or temporarily out of service. This operation can also be used for diagnostic or service operations. An H-bridge  61  of a type known in the art is connected to terminals  49  through diodes  71 ,  72 . A dc source  70  is connected across terminals  49  to supply power to the H-bridge  61 . Switching signals from a microelectronic processor (not shown) are provided to the H-bridge at terminals  62  and also to the power switch  40  through the gate switch  20 . The output of the H-bridge is fed to the converter  42  through connections  63 . From the converter  42  power is fed back through the main inverter  10  the main power supply  80  and to any loads. 
     The reversed charge circuit of FIG. 17 provides the galvanic-isolation feature. The low voltage side is coupled to the high-voltage side magnetically without a conduction link. All three secondary coils or just one secondary coil in converter  42  can be connected to the output of the H-bridge  61  for the low-to-high-voltage conversion. However, in order to have a higher voltage of the reversed charge, as shown in FIG. 17 only a partial winding of the one secondary coil is connected to the output of the H-bridge for the needed transformer turns ratio. The gate switching signals of the H-bridge  61  are operated in synchronization with one of the power electronic switching devices  18  in the main inverter  10  and one power switch  44  in the fourth leg of the inverter circuit. The unused power electronic switches  13 - 17  in the main inverter are turned to non-conducting with their gate switches  25 - 29  in the blocking situation. 
     In summary, the above-described methods and circuits of the present invention can be used with induction, permanent-magnet, and synchronous machines, with the transformer functions shown in the above-described circuits being achieved by using transformers. 
     When used with induction motors and the 3-phase main windings of the induction motor are used as the primary windings of the transformers, the motors should have a properly skewed rotor that will not react to the air-gap flux of the winding components of the zero-sequence current. When the main windings of an induction motor are not used as the primary windings of the transformers, any skew that is purely selected for the motor performance may be used. The circuits use the zero-sequence switchings of the main power electronic switching devices of the 3-phase inverter. This enables the use of one additional leg for each low-voltage control (see FIGS.  4  and  5 ). It is possible to use only one additional power-switching device in an additional leg (see FIG. 5) to control the low dc voltage output. 
     The single-phase connection circuit (see FIGS. 15 and 16) that shares components of the 3-phase inverter is suitable for the applications having a high duty-cycle tolerance of the low-voltage output. 
     The three-phase connections (see FIGS. 9,  10 ,  13 , and  14 ) are particularly suitable (see FIGS. 11 and 12) when used with two-phase modulation (i.e., allowing only four of the six power switching devices of the inverter being switched at any instant) (see FIGS.  7  and  8 ). This may be helpful with inverters whose low-voltage DC outputs may have a duty cycle problem when operating at a single-phase condition. 
     The low-to-high-voltage conversion can be achieved by using the additional low voltage H-bridge that is in synchronization with the corresponding power switching devices in the main inverter and the transformer function of the above circuits. Either all of the secondary coils, or one or two coils, or a partial winding of one coil can be connected to the H-bridge output for the needed transformer turns ratio. (An example of the low-to-high voltage is shown in FIG. 17.) 
     This has been a description of detailed examples of the invention. It will apparent to those of ordinary skill in the art that certain modifications might be made without departing from the scope of the invention, which is defined by the following claims.