Abstract:
A multi-channel transmission system operates by transmitting digitally-coded signals modulated in sequential symbol blocks at a plurality of carrier frequencies, and receiving and demodulating the digitally-coded signals, wherein, in the transmitting, a test signal is generated in at least one symbol block, the test signal containing at least one periodically continued, differentially coded, self-orthogonal sequence of a constant amplitude modulated to a part of the plurality of carrier frequencies within a symbol block in differential coding.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The invention relates to a method and an arrangement a transmission method and a multi-channel transmission system. 
     The invention is used in a COFDM (Coded Orthogonal Frequency Division Multiplexing) transmission system that has been proposed for digital radio. 
     2. Background Information 
     COFDM is a digital multi-channel modulation method. In such a method, the data signal to be transmitted is divided to a number of N (e.g. several 100) subchannels which lie next to one another in the frequency domain, with their spectra possibly also overlapping. With this division, the data rate transmitted in each subchannel is only a fraction of the original. The symbol duration is extended in the same ratio which has an advantageous effect if echoes appear on the transmission path. By selecting N to be sufficiently high, it is always possible for the symbol duration to be long relative to the maximum echo delay. Thus the symbol interference caused by echoes is reduced to the extent that the signal can be demodulated without distortion. 
     The COFDM transmission signal s(t) can be represented in the base band as a superposition of time and frequency shifted basic pulses b(t): ##EQU1## The basic pulse is here given by ##EQU2## j=√-1 is the imaginary unit. 
     The summation index i represents the symbol clock, index k represents the subchannel. The following parameters describe the COFDM modulation method: 
     T s  : symbol duration 
     t s  : utilized symbol duration 
     t g  : protection period 
     F s  : subchannel spacing 
     They are related by way of the equations T s  =t s  +t g  and F s  =1/t s . The quotient of the utilized symbol duration and the symbol duration, γ=t s  /T s  can be defined as a further parameter. In the COFDM variations presently being discussed for use in digital radio, it always applies that γ=0.8. Under consideration of these relationships, the COFDM system has only one free parameter from which the remaining can be derived. 
     The information to be transmitted is coded in complex symbols d i ,k. In COFDM, 4-phase keying is employed as the modulation method; it therefore applies that d i ,k ε(1, j, -1, -j). In order for the transmission to be insensitive to channel specific phase shifts, it is not the d i ,k symbols that are transmitted but the transmission signals s i ,k produced by differential coding s i ,k =s i-1 ,k ·d i ,k. Sometimes it is practical to combine the transmission symbols of all subchannels that were transmitted during the same time slot i into an N-dimensional vector s i . Such a vector is called a symbol block. 
     The generation of the COFDM transmission signal is effected, for example, digitally with the aid of the inverse fast Fourier transformation (IFFT). The block circuit diagram of a COFDM transmitter is shown in FIG. 14a. An IFFT is calculated for each time slot i. The output signal of the IFFT has the duration t s . It is continued periodically to become a signal of the duration T s . 
     The COFDM demodulator serves to recover the information carrying symbols d i ,k. For this purpose, the following values are formed from the receiver input signal r(t): ##EQU3## From this value, estimated values d i ,k are derived for the data symbols by differential demodulation d i ,k =r i ,k r* i-1 ,k, where r* is the conjugate complex to r. 
     The COFDM demodulator is also realizable digitally with the aid of the fast Fourier transformation (FFT). It is shown in FIG. 14b. A section of the duration t s  of the received signal is evaluated for every time slot. One section of the duration t g  remains unevaluated. The echoes of the signal from the preceding time slot fall into this section. 
     Data transmission in the COFDM system is frame oriented. A frame is a structured arrangement of timely successive symbol blocks. It has the following structure shown in FIG. 15: 
     The first symbol block s 1  in the frame is the zero symbol. It is characterized by the fact that no transmission signal is propagated. By means of an envelope detector, the receiver is able to detect the break in the field intensity. The distance between the zero symbols of the n th  and the (n+1) th  frame serves to synchronize the frames, the duration of the zero symbol serves to synchronize the symbols. The zero symbol is not processed by means of the FFT. It generally has a length other than T s . 
     The second symbol block s 2  in the frame is the phase reference symbol. It is required to initialize the differential demodulator. It is a complex sweep signal s 2 ,k =exp (j π k 2  /N). 
     The remainder of the frame is composed of information carrying symbol blocks. 
     COFDM requires that the carrier frequencies of transmitter and receiver match very precisely. The maximum tolerated deviation lies in an order of magnitude of 5% of the subchannel spacing F s . This can be realized only with very expensive special oscillators which are not suitable for mass production. It is better to employ a controlled oscillator in the receiver. At the moment of turn-on, however, this oscillator may have a frequency deviation in the order of magnitude of several subchannel spacings. 
     SUMMARY OF THE INVENTION 
     It is the object of the invention to measure the frequency deviation between transmitter and receiver by observing the received signal to then suitably correct the oscillator frequency in order to be able to use a control process. 
     This is accomplished by a system and method which includes transmitting digitally-coded signals modulated in sequential symbol blocks at a plurality of carrier frequencies, and receiving and demodulating the digitally-coded signals, wherein, in the transmitting, a test signal is generated in at least one symbol block, the test signal containing at least one periodically continued, differentially coded, self-orthogonal sequence of a constant amplitude modulated to a part of the plurality of carrier frequencies within a symbol block in differential coding. Advantageous features and/or modifications are defined below. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The above and other features of the invention will become apparent from the following detailed description taken with the drawings in which: 
     FIG. 1 shows a processing sequence schematically; 
     FIG. 2 shows the arrangement of an individual component group; 
     FIG. 3 shows a simplified processing sequence for a special case; 
     FIG. 4 shows an arrangement corresponding to the simplified processing sequence of FIG. 3; 
     FIG. 5 shows a sequence for the precise measurement of the frequency deviation; 
     FIG. 6 shows an arrangement corresponding to the FIG. 5 sequence; 
     FIG. 7 shows a special case processing sequence; 
     FIG. 8 shows an exemplary arrangement for the special case of FIG. 7; 
     FIG. 9 shows a sequence for measuring a channel pulse response; 
     FIG. 10 shows an exemplary configuration of an isolator; 
     FIG. 11 shows the configuration of an exemplary modified isolator; 
     FIGS. 12 and 13 show the configuration of an exemplary correlator and complex vector adder, respectively; 
     FIG. 14a shows a block circuit diagram of a COFDM transmitter which generates a transmission signal digitally using the inverse fast Fourier transformation (IFFT); 
     FIG. 14b shows a block circuit diagram of a COFDM demodulator realized digitally using the fast Fourier transformation (FFT); and 
     FIG. 15 shows a frame diagram for data transmission in the COFDM system. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The solution according to the invention resides in the periodically repeated transmission of a test signal with the aid of which the receiver is able to determine an existing frequency deviation. For practical reasons, it is advisable to transmit this test signal in each frame. The basic concept of the invention is to replace the previously existing sweep signal by this new test signal. 
     One advantage of the invention is that, with a single test signal, it is possible to measure the frequency deviation as well as, by determining the channel pulse response, the fine synchronization. 
     Another advantage is that the COFDM system requires no additional capacitance for this test signal since it replaces the previous phase reference signal. 
     The frequency deviation can advantageously be determined precisely and robustly. Thanks to the differential coding of the CAZAC sequence, the accuracy of the measured frequency deviation is not influenced by inaccuracies in the synchronization. 
     Since the test signal serves to simultaneously control time, frequency and phase, it is called the time/frequency/phase control symbol block (TFPC). 
     The TFPC is based on self-orthogonal sequences of a constant amplitude, so-called CAZAC (constant amplitude zero autocorrelation) sequences. These are finite sequences whose cyclic autocorrelation for mutually shifted sequences is zero. At least one CAZAC sequence is required to construct the TFPC. There may also be several, and one sequence may be employed several times. Therefore, a multitude of different combinations are suitable as TFPC. For that reason, the TFPC is initially defined generally and then clarified with the aid of an embodiment. 
     In one embodiment, the TFPC is the COFDM symbol block s 2 ,k. It is characterized in that at least one contiguous section s 2 ,ko, s 2 ,ko+1, . . . , s 2 ,ko+L-1 of the length L exists which, except for a complex multiplier A, corresponds to a finite sequence x i  (i=0, 1, . . . , L-1): 
     
         s.sub.2,ko+i =Ax.sub.i (i=0, 1, . . . , L-1). 
    
     Such a section is called a training sequence. 
     This allows for the generation of x i  as follows: 
     A four-value CAZAC sequence c m , (m=0, 1, . . . , M-1) of a length M&lt;L is continued periodically to reach the length L-1. 
     Then x i  is defined as follows: ##EQU4## χ i  is thus a differentially coded, periodically continued four-value CAZAC sequence. 
     The following limitations apply for the values M and L: 
     M must be the valid length of a four-value CAZAC sequence. At present, four-value CAZAC sequences are known to have the lengths 4, 8 and 16. It is also known that there are no four-value CAZAC sequences of other lengths less than 16. 
     In order for the correlation measurement to be described below to have precisely one distinct maximum, the relationship of L&lt;2M must be adhered to. 
     For the measuring range of the arrangement to be described below, that is, the maximum frequency deviation ΔF between transmitter and receiver that can be measured, the following applies: ##EQU5## 
     The divisions must here be understood as integer divisions. 
     It is favorable for L-M to be an odd number since the system then has the same measuring range as if a training sequence of the length L+1 were used. This is assumed to be the case in the discussions below. If L-M is even, a few algorithms must be modified, which is easily done. 
     The TFPC may contain a plurality of the above-described training sequences which may also overlap. The different training sequences may here be based on the same or also on different CAZAC sequences. 
     If symbols s 2 ,k of the TFPC exist which do not belong to at least one training sequence of the above-discussed type, these symbols may take on any desired values other than zero. 
     Particularly favorable conditions exist if all symbols s 2 ,k of the TFPC have the same amplitude. 
     In one embodiment, the TFPC is composed of a number of training sequences that is divisible by four. The TFPC is constructed of four CAZAC sequences produced from one CAZAC sequence by multiplication with 1, j, -1, -j. This arrangement results in particularly good accuracy and simultaneously particularly low computation efforts for the evaluation. The modulator employs a 128-point IFFT so that a total of 128 subchannels are available. They are numbered consecutively from 0 to 127. For filter technology reasons, only 86 of these 128 carriers are modulated, the so-called active subchannels. These are the carriers numbered 20 to 63 and 65 to 107. For technical reasons, the carrier numbered in this example cannot be utilized. 
     The basic CAZAC sequence is here given as follows: 
     
         __________________________________________________________________________i  0 1  2 3 4  5 6 7  8 9 10                       11                         12                           13                             14                               15__________________________________________________________________________C.sub.0,i   -j-1 1 1 -1 -j            1 -1 j 1 1 1 1 j 1 -1__________________________________________________________________________ 
    
     By multiplication with j, -1, -j, the following three CAZAC sequences are produced: 
     
         __________________________________________________________________________i  0  1 2  3  4 5  6  7 8  9  10 11 12 13 14 15__________________________________________________________________________C.sub.1,i   1  -j   j  j  -j           1  j  -j                   -1 j  j  j  j  -1 j  -jC.sub.2,i   j  1 -1 -1 1 j  -1 1 -j -1 -1 -1 -1 -j -1 1C.sub.3,i   -1 j -j -j j -1 -j j 1  -j -j -j -j 1  -j j__________________________________________________________________________ 
    
     The length of the CAZAC sequences in this example is M=16. L is selected to be 23; then a frequency deviation of ±3 subchannel spacings can be measured. 
     The CAZAC sequences c l ,i are extended by 3 elements each at the front and at the back and are differentially coded to yield the training sequences x l ,i. 
     
         __________________________________________________________________________i  0  1  2 3  4  5  6  7  8  9  10                             11__________________________________________________________________________x.sub.0,i   1  j  j -j -1 1  1  1  -1 j  j -jx.sub.1,i   1  -1 -j      -1 -1 j  -1 -j -1 -1 -j                             -1x.sub.2,i   1  -j j j  -1 -1 1  -1 -1 -j j jx.sub.3,i   1  1  -j      1  -1 -j -1 j  -1 1  -j                             1__________________________________________________________________________i 12 13 14  15 16 17 18 19  20 21 22__________________________________________________________________________x.sub.0,i  1  1  1   1  1  j  j  -j  -1 1  1x.sub.1,i  1  j  -1  -j 1  -1 -j -1  -1 j  -1x.sub.2,i  1  -1 1   -1 1  -j j  j   -1 -1 1x.sub. 3,i  1  -j -1  j  1  1  -j 1   -1 -j -1__________________________________________________________________________ 
    
     The following values are counted for the complex scaling factors A l  : 
     
         ______________________________________l          0     1           2   3______________________________________A.sub.l    1     1           -j  -1______________________________________ 
    
     Index offsets k l  are defined: 
     
         ______________________________________l          0     1            2   3______________________________________k.sub.l    21    40           65  84______________________________________ 
    
     The TFPC is composed of four training sequences. The TFPC vector of dimension 128 is here defined as follows: ##EQU6## 
     It must be noted, firstly, that in this example the training sequences overlap (in regions k=40 . . . 43 and k=84 . . . 87); secondly that subchannels K=63 and k=107 are not part of a training sequence and therefore were set arbitrarily; thirdly, that this TFPC has a constant amplitude in all active subchannels; and fourthly that it is composed of four CAZAC sequences which were produced by multiplication with 1, j, -1, -j from one CAZAC sequence. 
     For the evaluation of the TFPC it is assumed that the receiver is already roughly symbol synchronized on the basis of the evaluation of the zero symbol. The permissible synchronization error lies in an order of magnitude of ±0.5 t g . The receiver is then able to localize the TFPC signal in time and to subject it to an FFT. The vector r 2  according to Equation 3 is then present at the output of the FFT. This vector is now subjected to special processing which will be described below. 
     The receiver has stored the transmitted TFPC symbol. The training sequences contained in the TFPC, their position within the TFPC and the CAZAC sequences on which they are based are also known. 
     The processing of the TFPC is subdivided into several sub-tasks which in turn are composed of different processing steps. The sub-tasks are the following: 
     Rough measurement of the frequency deviation between transmitter and receiver. 
     Fine measurement of the frequency deviation between transmitter and receiver. 
     Measurement of the pulse response of the radio channel. 
     Buildup of the phase reference for the differential demodulation. 
     Common input value for all sub-tasks is γ 2 . 
     The rough measurement of the frequency deviation between transmitter and receiver is accurate to the order of magnitude of a subchannel distance. The following processing steps are required for this purpose: 
     1. Isolation of the received training sequences Υ 2 ,k0+1, The position of the training sequences in the transmitted TFPC is assumed to be known. A training sequence begins in the transmitted TFPC at s 2 ,k0. Then the (M+1)-dimensional vector u is calculated as follows: ##EQU7## If the TFPC contains several training sequences, the procedure is the same for each one of them and a plurality of vectors u i  are obtained. 
     2. Differential demodulation. The M-dimensional vector v is calculated with the aid of the following equation: 
     
         υ.sub.k =u.sub.k+1 u.sub.k * 0≦k&lt;M          (6) 
    
     If the TFPC contains several training sequences, the procedure is the same for each one of them and a plurality of vectors υ i  are obtained. 
     3. Cyclic correlation with the CAZAC sequence. Vector υ is correlated with the CAZAC sequence on which the training sequence is based. The (L-M)-dimensional vector w is calculated with the following equation: ##EQU8## If there are several training sequences in the TFPC, this calculation is made for each υ i  and one obtains the associated w i . These are then added to form a vector: w=Σ i  w i . 
     4. The w k  of the greatest amount is determined. The associated index k max  provides the searched-for frequency deviation: ΔF=k max  F s . 
     The correlation with the CAZAC sequence is simplified in that c m  takes on only the values 1, j, -1 and -j. 
     The above-described processing sequence is shown schematically in FIG. 1. The arrangement of the individual component groups is shown in FIG. 2. The vector r generated at the start of FFT processing for the symbol block containing the test signal is placed in a memory for intermediate storage for the purpose of further processing. This vector possesses the components r 0  through r N-1 . Test sequences are contained in symbol block i. These different test sequences are isolated separately from vector r as vectors U by isolators for different test sequences. For each test sequence, a vector V is formed in a differential demodulator; in a CAZAC correlator, this vector is cyclically correlated with the CAZAC sequence that forms the basis of the respective test sequence. During the correlation, the vectors w are generated for the different test sequences; these vectors are subsequently summed in a complex vector adder. The squared amount of the elements is formed in the sum vector, and the element having the maximum amount is determined. The index k max  associated with this element serves in the rough determination of the frequency deviation. The ascertained results can be stored in a further memory. 
     In the configuration for isolation and differential demodulation of training sequences shown in FIG. 10, the starting values of the individual training sequences within the majority of carrier frequencies are stored in a read-only memory (ROM). These values are converted into control addresses in an address generator in order to actuate the memory for the output vectors of the FFT. The different training sequences can be isolated in a simple manner through the entry of the starting addresses for the training sequences. The stored values associated with a training sequence are read out in a predetermined reading cycle as of the starting address, and a series of real components and a series of imaginary components are formed from the complex value sequence in a 1-to-2 demultiplexer and each entered into one of two shift registers (FIFO). 
     By means of a delay element for each register cycle, two complex values that are successive in the training sequence are always available simultaneously in the signal paths of real components and imaginary components. 
     A further complex value series which is fed to the correlator via further shift registers results from complex multiplication of two such complex values according to Equation (6). 
     FIG. 12 shows a correlator for four-value CAZAC sequences. The correlator is represented in a standard manner as a shift register having a plurality of register stages which are delayed with respect to one another by one register cycle. In this correlator, the outputs of the individual register stages are weighted with values of the CAZAC sequence forming the basis of the training sequence and summed. In a four-value CAZAC sequence, only the weightings 1, j, -1, -j are provided, so the correlation is significantly simplified by the insertion of two inverters. In addition to real component Re and imaginary component Im, for each register cycle, the inverted components -Re and -Im are also present for a complex value. In place of the multiplication in weighting, one of these four values can simply be selected, as indicated by the solid lines in the drawing. The correlator outputs are fed to the complex vector adder via further shift registers. 
     The complex vector addition, as shown in FIG. 13, simply provides the summation of a plurality of output vectors of the correlators separated according to real components and imaginary components for different test sequences, resulting in a complex sum vector. From this vector, a squared-amount vector can be formed by squaring the real parts and imaginary parts of the individual vector elements; within this vector, the element having the maximum amount is determined. The results of the maximum search and the complex vector addition are stored in a result memory. The method according to the invention is considerably simplified if the TFPC is constructed of four CAZAC sequences which were produced from one CAZAC sequence. Then the vectors belonging to the different training sequences can be combined directly already after the differential demodulation. For this purpose u=Σ i  α i  u i  is formed. The α i  values then take on the values 1, j, -1 and -j. α i  is a conjugate complex to the factor with which the original CAZAC sequence was multiplied to obtain the CAZAC sequence that is the basis of the i th  training sequence. The combination of the signals now requires the calculation of the correlation with the CAZAC sequence only once. FIG. 3 shows the simplified processing sequence in this special case. FIG. 4 shows the corresponding arrangement. The configuration and processing are effected separately until the complex vectors V i  are formed through correlation, as in different training sequences. The simplification arises from the fact that the complex vector addition is already applied to the output vectors of the differential demodulators, and only one sum vector formed during this is subjected to correlation. The explanations for FIGS. 1 and 2 apply to the function. 
     The method for precisely determining the frequency deviation between transmitter and receiver makes it possible to determine this deviation with an accuracy to a fraction of the subcarrier spacing. It assumes a rough knowledge of the frequency deviation, that is, it builds on the above-described method of roughly determining the frequency deviation. There the index k max  of the correlation maximum was found. This index is assumed to be known for the processing to follow. It is also assumed that vector w is known. Then the precise determination of the frequency deviation is effected as follows: 
     1. Isolation of the training sequences. The position of the training sequences in the transmitted TFPC is assumed to be known. A training sequence begins in the transmitted TFPC at s 2 ,k0. Then the (M+2)-dimensional vector u&#39; is calculated as follows: ##EQU9## If the TFPC contains several training sequences, the procedure is the same for each one of them and several vectors u i  &#39; are obtained. 
     2. Modified differential demodulation. The M-dimensional vector υ&#39; is determined with the aid of the following equation: 
     
         υ.sub.k &#39;=u.sub.k+1 &#39;u.sub.k+1 &#39;*+u.sub.k+2.sup.&#39;u.sub.k &#39;* 0≦k&lt;M                                              (9) 
    
     If the TFPC contains several training sequences, the procedure is the same for each one of them and several vectors υ i  &#39; are obtained. 
     3. Reduced cyclical correlation with the CAZAC sequence. Vector υ&#39; is correlated with the CAZAC sequence on which the training sequence is based. The following values are calculated: ##EQU10## If there are several training sequences in the TFPC, this calculation is performed for each υ i  &#39; and the associated B i  and C i  are obtained. These are then added together in each case: C=Σ i  C i  and B=Σ i  B i . 
     4. From the rough determination of the frequency deviation the value w kmax  is available. The values D=2 W kmax  -C are calculated. The precise measurement of the frequency deviation is then effected as follows: ##EQU11## 
     The above-described sequence for the precise measurement of the frequency deviation is shown in FIG. 5. FIG. 6 shows a corresponding arrangement. FIG. 11 shows the configuration of an exemplary, modified isolator. 
     The modified isolator and demodulator shown in FIG. 11 is configured similarly to the one shown in FIG. 10, but differs from it in the following essential points: 
     a) the shift k max  ascertained from the rough determination of the frequency deviation is superposed onto the starting value for a training sequence stored in the read-only memory (ROM), so the segment of the receiver vector that is shifted by k max  is read out of the FFT output memory as a training sequence; 
     b) in differential modulation, the values multiplied together in a complex manner corresponding to the second sum in Equation (9) are two register cycles apart; 
     c) for the complex value between the two values processed according to b), the squared amount is formed corresponding to the first sum in Equation (9) and added to the complex product according to b) in accordance with Equation (9); since the squared amount is always real, this addition only takes place in the signal path of the real part. 
     With a TFPC constructed of four CAZAC sequences it is possible to combine the various training sequences before the cyclic correlation. The reduction in computation effort realized doing this, however, since only two values are calculated for the correlation function, is not so significant as in the rough determination of the frequency deviation where L-M values were calculated. FIG. 7 shows the sequence in this special case. FIG. 8 depicts an exemplary arrangement for the special case. The differences between the example shown in FIGS. 7 and 8 and the one shown in FIGS. 5 and 6 lie in the simplification, namely that the complex vector addition is already applied to the output vectors of the differential demodulators, and only one sum vector formed during this is subjected to correlation. Refer to the explanations of FIGS. 5 and 6 for the function. 
     The sequence for measuring the channel pulse response is shown in FIG. 9. 
     To measure the channel pulse response, vector r 2  is multiplied element by element with the conjugate complex TFPC transmission signal s 2  which is stored in the receiver. This results in the N-dimensional vector H as follows: 
     
         H.sub.k =r.sub.2,k s*.sub.2,k                              (13) 
    
     H is an estimate of the transmission function of the Channel. This vector is transformed with the inverse, fast Fourier transformation (IFFT) into the vector h which constitutes an estimate of the channel pulse response. Fine synchronization is effected by determining the first index k sync  at which the channel pulse response has a significant energy component. 
     The TFPC is also suitable as a phase reference since it has an energy content other than zero in every active subchannel. The TFPC that has been normalized in each subchannel to an amplitude of 1 is fed, at the transmitter, to the differential coder and, at the receiver, to the differential demodulator. If the TFPC has constant energy components in all subchannels, this is an advantage since the phase noise in the noise channel is then the same in all subchannels.