Abstract:
Harmonic techniques are employed to leverage low-cost, ordinary surface mount technology (SMT) to high microwave frequencies where tight beamforming with a small antenna makes reliable, high-accuracy pulse-echo radar systems possible. The implementation comprises a 24 GHz short-pulse transceiver comprised of a pulsed harmonic oscillator employed as a transmitter and an integrating, pulsed harmonic sampler employed as a receiver. The transmit oscillator generates a very short (0.5 ns) phase-coherent harmonic-rich oscillation at a sub-multiple of the actual transmitter frequency. A receiver local oscillator operates at a sub-multiple of the transmit frequency and is triggered with controlled timing to provide a very short (0.5 ns), phase-coherent local oscillator burst. The local oscillator burst is coupled to an integrating harmonic sampler to produce an integrated, equivalent-time replica of the received RF. The harmonic techniques overcome four major problems with non-harmonic approaches: 1) expensive, precision assembly, 2) high local oscillator noise, 3) sluggish oscillator startup, and 4) spurious local oscillator injection locking on external RF. The transceiver can be used for automotive backup and collision warning, precision radar rangefinding for fluid level sensing and robotics, precision radiolocation, wideband communications, and time-resolved holographic imaging.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to wide bandwidth pulsed microwave transmitters and receivers, and more particularly to short-range, sub-nanosecond pulse, phase-coherent K-band radars. 
     2. Description of Related Art 
     Range measurement of close-range targets is of great interest to a number of industries. Automotive backup warning radar, fluid level sensing in tanks and vats, material level sensing in silos, safety systems, home “do-it-yourself” projects, and aids to the blind are but a few of the applications for short-range non-contact range measurement. Radar range measurement appears to be the technology of choice but has yet to make significant penetration into these markets. The main shortcoming with radar has been the difficulty in realizing a low-cost short-pulse radar with a narrow antenna beam. 
     A potentially suitable radar is ultra-wideband (UWB) radar, for example, ground penetrating radar as disclosed in U.S. Pat. No. 4,698,634 by Alongi or Micropower Impulse Radar (MIR) in U.S. Pat. No. 5,774,091 by McEwan. UWB radar emits suitably short pulses (&lt;Ins) but has serious drawbacks; its low frequency spectrum can create interference with countless other spectrum users below 3 GHz, and its low frequency spectrum (i.e., long wavelength) prevents narrow antenna beam formation with a compact antenna. 
     Ultrasound is a potential technology that is both simple and inexpensive. Unfortunately, it is of limited accuracy since the speed of sound varies 10% over outdoor temperatures. Accuracy is of central importance in tank level measurements and construction applications, and 10% accuracy is simply not consistent with modern requirements. Accuracies of 1% to 0.01% are needed. These accuracies can be met with pulse-echo radar using precision timing techniques as will be described herein. 
     In addition to limited accuracy, ultrasound is susceptible to extraneous acoustic noise, and water or dirt overcoatings on its transducers can disable it. In spite of these limitations, ultrasound has been a popular ranging technology due to its simplicity and its ability to form a narrow beam with a small transducer. A narrow beam is needed to reduce clutter reflections from off-axis objects, such as a tank wall. A narrow beam also implies high antenna gain, which improves signal to noise (S/N) ratio. 
     While both limited and antiquated, ultrasonic rangefinding remains the dominant non-contact range measurement technology since there have been no real alternatives. One might consider an optical approach to rangefinding, such as a laser rangefinder or a video system. However, optical systems also lack environmental ruggedness—the optics cannot be located behind a decorative panel and can be disabled by an overcoating of water, snow, ice or dirt. Clearly, a better technology is needed. 
     Radar rangefinders are environmentally rugged: the speed of light (at which radar waves travel) does not vary with temperature (for all practical purposes), and radar waves propagate freely through wood walls, gypsum walls and plastic panels, even with an overcoating of water, ice, snow or dirt. 
     Pulse-echo radars operating in the 24 GHz band have a wavelength of 12.5 mm, which is almost exactly the same wavelength as 25 KHz ultrasound. Since antenna beamwidth is determined by the wavelength to antenna aperture ratio, radar and ultrasound will have comparably narrow beamwidths with the same antenna/transducer footprint. 
     An ultrasonic rangefinder may typically transmit a burst of 12 sinusoidal cycles of acoustic energy with a corresponding pulse width that defines the two-object resolution of the system. Of course, its incremental resolution is not a function of emitted pulse width, but that of the timing system. A 24 GHz radar with the same two-object resolution as the 12-cycle ultrasound system needs to transmit a 12 cycle, 0.5-nanosecond sinusoidal burst at 24 GHz, since the wavelengths are comparable. Clearly, the radar needs to have a wide bandwidth, on the order of 1-2 GHz. 
     Prior art pulse echo radars do not exhibit the combination of 1) K-band RF operation, e.g., 24 GHz, 2) sub-nanosecond RF pulse width, 3) extreme phase coherence (&lt;10-picoseconds for the entire transmit-receive system, 4) expanded time output with ultrasonic parameters, 5) simple assembly with low cost surface mount technology (SMT) components, and 6) commercially appealing size and cost. Clearly, a new technology is needed. 
     Attempts by the present inventor to develop a 24 GHz radar rangefinder using SMT components were met with frustration and failure—a quarter wavelength at 24 GHz is 3 mm or even less when material dielectric constants are included. Since SMT components have dimensions on the order of 3 mm, wavelength effects are a severe limitation. 
     One approach to counter the effect of diminishing wavelength is to decrease component size with monolithic technology such as GaAs MMIC (monolithic microwave integrated circuit). Unfortunately, the high cost of GaAs MMIC, about $10 per chip, puts radar in an uncompetitive position relative to ultrasound, which can be fully implemented on a single low cost silicon chip. A pulse-echo radar system with transmit and receive MMICs, and support circuitry might cost $50 to manufacture, after factoring-in expensive assembly techniques for very small high bandwidth components and special circuit board materials. In contrast, a complete ultrasound system can be manufactured for under $5. 
     SUMMARY OF THE INVENTION 
     One solution to the cost problem of a radar rangefinder is to employ SMT components with as few microwave semiconductors as possible. To implement this approach, unique wide-bandwidth harmonic techniques were developed for the present invention. Accordingly, a pulsed transmit oscillator operates at a sub-multiple of a transmit frequency and a strong harmonic is extracted for transmission. Similarly, a pulsed local oscillator in the receiver operates at one-half or one-quarter the transmit frequency and drives a harmonic sampler operating at the transmit frequency. Thus, all the critical microwave components operate at frequencies where SMT components are viable, typically at less than 15 GHz. 
     In a typical radar configuration, the transmit oscillator is connected to an antenna which radiates a short RF burst at a harmonic frequency. Echo bursts are received by a receive antenna and sampled by a harmonic sampler that is driven by a sub-harmonic (to the radiated frequency) RF burst. The timing of the RF burst is slowly swept to produce an equivalent time analog replica of the received echo burst, which can be used to determine target characteristics such as size and range. 
     In a preferred embodiment, the transmitter of the present invention uses a single, pulsed GaAsFET transistor operating at 12 GHz and frequency doubled to 24 GHz using a resonant antenna and a waveguide beyond cutoff to extract the desired second harmonic. The GaAsFET is the same as that used by “Dish” TV systems, so its cost has been driven down by this popular consumer electronics technology to about $1 in volume. 
     The receiver in the preferred embodiment uses a silicon transistor operating in a short pulse mode at 6 GHz and effectively frequency quadrupled to 24 GHz by a harmonic sampler. The silicon transistor cost is 15 cents in volume. The only other microwave semiconductor component in the system is the detector diode. The present invention uniquely employs a simple technique to double the bandwidth of commercially available SMT detector diodes, thereby allowing the use of a 70-cent detector diode. With an RF lineup costing under $2, it is quite feasible to manufacture a complete 24 GHz rangefinder for under $5, or about ten times lower than the GaAs MMIC approach. 
     The present invention emits a short sinusoidal RF burst containing a limited number of cycles, such as 12 RF cycles. Thus, there is a need to generate 500-picosecond wide RF bursts at 24 GHz. As a further constraint, the sinusoidal cycles within the RF burst must be phase coherent with the timing pulses that trigger the burst, i.e., the trigger jitter must be less than one-quarter of an RF cycle (10-picoseconds), and ideally on the order of 1-picosecond. The technology to do this cannot be found in the prior art. 
     To simplify signal processing and to make the entire rangefinder practical, the present invention employs expanded time techniques, also known as equivalent time (ET). ET is a beat-frequency effect produced by sampling echoes at a slightly slower rate than the transmitted pulse rate. The net effect is very similar to shining a strobe light on a fan blade, and adjusting the strobe frequency so the blade appears to rotate very slowly. By this analogy, the rapidly rotating fan blade represents the realtime pulses travelling at the speed of light, the strobe is an electronic gate in the receiver (or the pulsed local oscillator and harmonic sampler described herein), and the slowly rotating visual effect is the expanded time radar video output. “Video” is used here in the common radar parlance, and is not to be confused with television or visual signals. 
     An ET pulse-echo radar system transmits pulses, and after a delay its receiver is gated at a particular point in time, or equivalently, in range. The timing of the gate is typically swept across a range of delays (e.g., 0-100 ns) in a matter of milliseconds, such that the receiver video output is a scan-like waveform which replicates events occurring on a realtime 0-100 ns scale by an equivalent time millisecond-scale. Equivalent time techniques are commonly used in wideband sampling oscilloscopes and will not be dwelt upon here. What is novel in this invention is the use of a very short (e.g., 0.5 ns) sub-harmonic microwave RF burst as the sampling gate, particularly where the microwave burst is a sub-harmonic of the received RF pulse. As a further enhancement, the sampling gate, or harmonic sampler of the present invention, coherently integrates multiple repetitions of the received microwave RF pulses. 
     Precision timing circuits are required for accurate equivalent time systems. Timing circuits having scale factor accuracies on the order of several tens of picoseconds or better can be realized with a Delay Locked Loop (DLL) such as a “Precision Digital Pulse Phase Generator” as disclosed by McEwan in U.S. Pat. No. 5,563,605, or in copending application, “Phase-Comparator-Less Delay Locked Loop”, Ser. No. 09/084,541, by McEwan. Alternatively, dual crystal clocks, one for transmit and one for receive, can be employed, where the receive clock is locked to a small offset frequency from the transmit clock, such as 100 Hz, thereby causing a steady phase slip of one complete clock cycle  100  times per second. In the process, the receive sampler timing smoothly sweeps across one complete pulse repetition interval (e.g., a PRI=100 ns for a 10 MHz clock) every 10 ms in equivalent time. 
     The present invention is a precision radar rangefinder employing pulsed harmonic techniques and low cost manufacturing technology. It can be used in low cost radars for tank level measurements, including 0.01% accurate custody transfer measurements, industrial and robotic controls, vehicle backup warning and collision radars, and general rangefinding applications. While illustrated with reference to a radar rangefinder, the invention also encompasses other pulsed RF systems, including (1) precision time-of-flight and time-of-arrival radiolocation systems similar to U.S. Pat. Nos. 5,510,800 and 5,661,490, both by McEwan, (2) wideband pulsed-RF communication systems, and (3) holographic imaging systems. Since the present invention is phase coherent, microwave holograms can be formed using techniques known in the art, where the customary holographic reference beam is conveniently replaced by the internal phase coherent timing of the present invention. 
     A primary object of the present invention is to provide a precision, low cost radar ranging system with a narrow beamwidth using a small antenna. 
     Yet another object of the present invention is to provide a radar ranging system with an expanded-time video output signal that is similar to an ultrasonic rangefinder signal to facilitate low cost and low power signal processing. 
     Still another object of the present invention is to provide a K-band harmonic oscillator and harmonic sampler that can be fabricated with common surface mount components. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram showing the short pulse microwave transceiver of the present invention. 
     FIG. 2 is a schematic diagram of a pulse driver and harmonic oscillator of the present invention. 
     FIG. 3 a  shows the waveform at the antenna terminal of the harmonic oscillator of FIG.  2 . 
     FIG. 3 b  shows the radiated waveform of the harmonic oscillator of FIG.  2 . 
     FIG. 3 c  shows the radiated waveform of the harmonic oscillator of FIG. 2 with the RF pulse width adjusted to a minimum. 
     FIG. 3 d  shows the detected baseband video waveform of a pulse transmitted and received by the system of FIG. 1 using the transmitter of FIG.  2  and the receiver of FIG.  4 . The waveform represents a target reflection at 5-meters range. 
     FIG. 4 is a schematic diagram of a pulse driver, a GaAsFET oscillator, and a harmonic sampler of the present invention. 
     FIG. 5 is a schematic diagram of a pulse driver, a silicon bipolar transistor oscillator, and a harmonic sampler of the present invention. 
     FIG. 6 a  is a schematic diagram of a fast driver and short pulse fundamental oscillator of the present invention. 
     FIG. 6 b  is a physical layout diagram of the short pulse fundamental oscillator of FIG. 6 a.   
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     A detailed description of the present invention is provided below with reference to the figures. While illustrative component values and circuit parameters are given, other embodiments can be constructed with other component values and circuit parameters. All U.S. patents and copending U.S. applications cited herein are herein incorporated by reference. 
     FIG. 1 is a block diagram of a microwave transceiver  10  of the present invention. A harmonic oscillator  12  receives clock pulses from TX clock  14  via pulse driver  18  and produces RF burst pulses (transmit pulses) at the transmit antenna  16 . TX Clock  14  typically produces a 1-10 MHz squarewave that is passed through a pulse driver  18  to form ˜1 ns wide pulses with rise and fall times below 100 ps. Thus, the drive pulses are clock pulses with very fast rise and fall times. Therefore, the TX clock  14  and pulse driver  18  may together be viewed as a clock signal generator. Alternatively, if TX clock  14  produces fast rise and fall time pulses, a separate driver  18  is not required. These short pulses bias-on the harmonic oscillator, which designed to start and stop oscillating very rapidly as a function of applied bias. The oscillations are phase coherent with the drive pulses, i.e., the phase of the RF sinusoids relative to the drive pulse remains constant each time the oscillator is started—there is no significant clock-to-RF jitter. 
     A high degree of phase coherence can only be obtained with a very fast risetime drive pulse that shock excites the oscillator into oscillation. Accordingly, the pulse drivers of the present invention have risetimes of typically less than 100 ps. 
     Harmonic oscillator  12  typically operates at a fundamental frequency of 12.05 GHz with a second harmonic at 24.1 GHz. A frequency of 24.1 GHz or thereabouts is preferred since commercial and consumer devices such as radar rangefinders can operate in the 24.0-24.25 GHz band without a license. The transmitted RF bursts are typically 12 cycles long at a carrier frequency of 24.1 GHz. 
     In a less preferred mode, a fundamental frequency oscillator  20  operating at 24.1 GHz can be used instead of a harmonic oscillator  12 . A fundamental mode oscillator has a higher output amplitude but is far more difficult to implement in surface mount technology (SMT) since the operating quarter-wavelength at 24 GHz is 3 mm, about the same length as a SMT transistor package (including leads) that would be used for the oscillator. Obtaining oscillations in a SMT package at 24 GHz is nearly impossible. The present invention solves that problem in a fashion that will be discussed with reference to FIGS. 6 a  and  6   b  below. However, the required physical layout does not conform to common SMT design rules and is thus less desirable than a harmonic oscillator operating at 12 GHz where standard SMT layout rules can be adhered to. 
     A receive (RX) clock  22  typically produces a 1-10 MHz squarewave that is passed through a pulse driver  24  to form ˜1 ns wide pulses with rise and fall times below 100 ps. These short pulses bias-on the local oscillator  26 , which is designed to commence oscillation very rapidly. The local oscillator operates at a sub-harmonic of 24.1 GHz, i.e., 12.05 GHz or 6.025 GHz. The 0.5 ns wide RF bursts generated by the local oscillator gate the harmonic sampler  30  to form a detected signal that is amplified by a low frequency amplifier  31  and filtered in bandpass filter  32  to produce a baseband video signal. 
     The RX clock  22  is typically swept in time relative to the TX clock  14 . Sweeping is realized by operating the RX clock  22  at a slightly lower, or offset, frequency than TX clock  14 . Typically, the TX clock is a first quartz crystal oscillator, and the RX clock is a second quartz crystal oscillator that is phase locked to a typical offset of 100 Hz from the TX clock, such that the phase of the RX clock slips one full pulse repetition interval (PRI) every 10 ms. In this case, timing control unit  34  performs a phase-lock function between the TX and RX oscillators. 
     Alternatively, a swept delay RX clock can be realized by the timing control unit  34  alone (acting as a delay element with controlled delay), when connected between TX clock  14  and pulse driver  24 , with the offset frequency RX clock  22  omitted and timing control unit  34  providing the RX clock. A means to accomplish precision swept timing has been described in co-pending application “Phase-Comparator-Less Delay Locked Loop,” Ser. No. 09/084,541, by McEwan, which exhibits 0.01% linearity across a 10-meter measurement range. 
     With either type of swept timing, an equivalent time (ET) replica of the transmitted RF signal radiated from antenna  16  and received by antenna  18  (or from a remote transmitter in the case of a radiolocation or communication system) appears at the harmonic sampler  30  output. The ET replica resembles the RF signal, except it occurs on a slow time scale such as 10 ms. FIG. 3 d  provides an example of an ET replica signal on a 2-millisecond/div scale. In the typical case where the local oscillator RF pulse width contains more than one RF cycle, the ET output is broadened somewhat by the convolution of the harmonic oscillator pulse and the local oscillator pulse. 
     The TX clock  14  can be modulated in frequency or phase, or with ON-OFF gating by optional modulator  35  for various purposes known in the art, such as spreading the spectral lines generated by harmonic oscillator  12  to reduce interference to other spectrum users. The modulation may be coded to distinguish the desired received signal from that of others. Modulator  36  is connected to the RX clock  22  or timing circuit  34  to demodulate coded signals from a remote transmitter using synchronization means know in the art of spread spectrum radio technology. 
     FIG. 2 is a detailed schematic of the pulse driver  18  and harmonic oscillator  12  (which form the transmitter) of the present invention. The TX clock signal is coupled through NOR gate  40  and emitter follower transistor  42  to provide a gate pulse  44  with a fast risetime. NOR gate  46  is coupled to NOR gate  40  to limit the pulse width from NOR gate  40 . The TX clock signal is also coupled through an adjustable RC delay network  48 , NOR gate  50 , and inverter transistor  52  to switch the gate pulse  44  to a low level (ground level) after a delay set by RC network  48 . RC network  48  can be adjusted to provide gate pulse widths from 0 to 10 ns, typically. 
     The particular setting of the gate pulse width depends on the RF parameters required of the RF system. Very short RF pulses are needed for high spatial resolution or for high data rate systems, whereas wide pulses allow for a lower bandwidth bandpass filter  32  (at the video output) and thus better signal to noise ratio. Also, the RF pulse width may be limited by regulatory constraints where an overly short pulse may have excessive bandwidth. 
     Harmonic oscillator  12  is comprised of a GaAsFET  54 , several microstrip resonators  56  (labeled L1) connected to the source and drain of GaAsFET  54 , and antenna  58  connected to the drain. The microstrip resonators  56  are all approximately one-quarter wavelength long at the fundamental frequency of oscillation, which is 12.05 GHz for a 24.1 GHz radiated frequency. Antenna  58  is one-quarter wavelength long at the second harmonic, or 24.1 GHz. Thus, it resonates and provides a low impedance to the drain of the GaAsFET for maximum harmonic extraction. Experiments show that second harmonic generation at 24.1 GHz is only 7 dB lower than the same transistor operating in the fundamental mode at a frequency of 24.1 GHz. 
     One of the microstrip resonators has a short wire or metal tab  64  attached to it to fine-tune the oscillator. During production the tab  64  can be bent to vary its capacitance to ground and thereby tune the RF frequency. Alternatively, a metal screw can be brought close to one of the microstrips for fine-tuning. 
     The entire oscillator circuit  12  is placed inside a waveguide  60  that operates beyond cutoff at 12 GHz to substantially reduce unwanted radiation at the 12 GHz fundamental frequency. Thus, waveguide  60  is somewhat greater than one-quarter wavelength high (so the quarter-wave antenna fits inside) and one-half wavelength wide at 24 GHz. A horn  62  is coupled to the waveguide to provide gain and reduce sidelobe radiation. In one prototype, the horn flares from 4×6 mm to 20×50 mm and provides about 18 dB gain at 24 GHz. 
     Harmonic oscillator  12  operates at 12 GHz (but produces a transmit signal at 24 GHz) and has three key advantages over a fundamental mode oscillator  20  (described with reference to FIGS. 6 a  and  6   b  below) operating at 24 GHz. First, all the microstrip dimensions are twice as long. This is critical, since the microstrip dimensions at 24 GHz become smaller than the SMT transistor leads. Second, the harmonic oscillator can start up very rapidly since the GaAsFET has much higher gain at 12 GHz than at 24 GHz. This is particularly true when package parasitics are factored in. There appears to be a “wall” at about 22 GHz, beyond which SMT devices will not oscillate without special layout techniques (as seen in FIG. 6 b ). 
     The third advantage to a harmonic oscillator is that the startup phase of oscillation is very independent of external RF at double frequency, i.e., received signals at 24 GHz. To illustrate this injection locking problem, experiments show that the slightest RF present at 24 GHz will alter both the startup phase and the oscillation envelope risetime of a 24 GHz oscillator. Consequently, vitally needed phase coherence is lost, and when the oscillator is used as a local oscillator, the modulated local oscillator envelope causes extremely high noise in the detector. 
     In contrast, external 24 GHz RF has essentially no effect on the same oscillator when operated at 12 GHz. This is a critical factor in the present invention, which makes use of a high degree of phase coherence between the drive pulse and the emitted and detected RF pulse. The typical jitter between the drive pulse and the transmitted RF pulse is on the order of 1 ps. If the jitter exceeds one-quarter of an RF cycle, or  1  Ops for a 24 GHz system, the detected signal will degrade substantially. This is due to the fact that the harmonic sampler detects and holds the received RF signal over more than one clock cycle. In other words, the transceiver coherently integrates more than one pulse, and multiple pulse integration of random phase pulses will integrate to zero, and not to the desired signal. This will be further explained with reference to FIG. 4 below. 
     GaAsFET transistor  54  is a NEC type NE42484A, silicon bipolar transistors  42  and  52  are NEC type NE68533, and NOR gates  40 ,  46 ,  50  are type 74AC02. 
     FIG. 3 a  shows the startup waveform of the harmonic oscillator of FIG.  2 . At the onset of oscillation, a fundamental frequency oscillation occurs at 12 GHz. This quickly breaks into 24 GHz oscillations as the oscillation progresses, even though measurements at the source of the GaAsFET indicate the transistor continues to oscillate at 12 GHz. The oscillation at the drain of the GaAsFET is dominated by the second harmonic at 24 GHz due to the attachment of a 24 GHz resonant antenna. 
     FIG. 3 b  shows the radiated waveform of the harmonic oscillator of FIG. 2, after exiting the waveguide. The waveguide filters out the 12 GHz fundamental. The radiated second harmonic at 24 GHz is about +3 dBm in amplitude. 
     FIG. 3 c  shows the waveform of the harmonic oscillator of FIG. 3 b  on a slower time scale (500 ps/div) and with the RC delay network  48  of FIG. 2 adjusted for a minimum RF pulse width, or about 0.5 ns. 
     FIG. 3 d  shows the detected video waveform of an echo from a metal plate at 5-meters range using the 24 GHz harmonic oscillator of FIG.  2  and the harmonic sampler of FIG.  4 . It is an equivalent time waveform on a 2-millisecond per division scale that represents an 800 ps/div real time scale. The RC delay network  48  was adjusted to provide a ˜2 ns RF pulse width from the harmonic oscillator of FIG.  2 . 
     FIG. 4 is a detailed schematic of the pulse driver  24 , local oscillator  26 , and a frequency doubling harmonic sampler  30  (which form the receiver) of the present invention. Pulse driver  24  is identical to pulse driver  18  of FIGS. 1 and 2. Local oscillator  26  is essentially the same oscillator  12  of FIGS. 1 and 2, except it has no antenna connection to the GaAsFET Q 3 . Instead a 12 GHz quarter-wave microstrip resonator  70  is coupled to a 24 Ghz quarter-wave microstrip resonator  72  in the harmonic sampler circuit  30 . Thus 12 GHz oscillations are provided to the harmonic sampler local oscillator port, i.e., microstrip  72 , and 24 GHz signals are provided at its RF input port  76  to which receiver antenna  74  is connected. Antenna  74  is a 24 GHz quarter-wave monopole located in a 6 mm-wide waveguide  77 . The waveguide has a horn  78  attached as described with reference to FIG.  2 . The waveguide cuts off below 24 GHz, thereby preventing interference from low frequency sources. 
     The microstrip resonators L1 connected to Q3 including coupling resonator  70 , are all approximately one-quarter wavelength long at the fundamental frequency of oscillation of Q3, i.e., 12 GHz. The microstrip resonators L2 connected to harmonic sampler  30  are all approximately one-quarter wavelength long at the received frequency, i.e., 24 GHz. A quarter-wave microstrip L 2  is connected to RF input port  76 , and operates as described in reference to FIG.  5 . 
     Sampler diodes  80  are connected as series pairs and then as back-to-back (or anti-parallel) pairs. The diodes are connected in series to cut the lead-to-lead capacitance in half. Yet the lead-to-lead inductance of two leads in a series connection remains about the same as two leads for a single diode. This assumes the diodes are on a common substrate and packaged in a single SMT package. The inductance between the diodes connected in series is limited to a small on-chip inductance and is negligible compared to the lead inductance of the package. Experiments show that a single 0.25 pF Schottky diode in a small SMT package has a detector bandwidth of about 10 GHz, whereas the series pair has a bandwidth of greater than 20 GHz, without matching. Schottky sampling diodes  80  are HP type HSMS-286C and the op amp  86  of low frequency amp  31  is a TI type TLO71. 
     The back-to-back connection of detector diodes  80  results in conduction on each half cycle of the 12 GHz oscillation provided by the local oscillator  26 . The net effect is the same as a single diode conducting on every full cycle of a 24 GHz local oscillator. Accordingly, the sampler  30  operates on the second harmonic of the local oscillator  26 . Experiments show that this frequency doubling harmonic sampler has nearly the same sensitivity as a non-harmonic 24 GHz local oscillator and sampler. 
     In addition to harmonic sampling, a key advantage to the back-to-back diode connection is rejection of local oscillator noise, since the back-to-back connection develops zero offset bias (in principle), and consequently there can be little noise due to the local oscillator. Tests show that the 12 GHz GaAsFET circuit of FIG. 4 can achieve the same noise performance as the 6 GHz silicon bipolar local oscillator of FIG. 5 below, when either is configured to harmonically sample at 24 GHz. The advantage of the GaAsFET is that it can generate a shorter RF burst than the silicon bipolar oscillator, and consequently, the 24 GHz harmonic sampler  30  will have wider bandwidth. 
     Quarter wave microstrip  82  provides an RF short at 24 GHz to the video output end of the sampler to prevent RF leakage out to the amplifier  31 . It also enhances the detection efficiency of diodes  80 . Amplifier  31  amplitude-scales and impedance-buffers the detected signal. The signal is then passed through a bandpass filter  32  to limit noise and, in some cases, to provide Doppler filtering for velocity discrimination. 
     FIG. 5 is a detailed schematic of a pulse driver  24 , a local oscillator  26 , and a frequency quadrupling harmonic sampler  30  of the present invention. The configuration of this pulse driver is different from that of FIGS. 2 and 4. An RX clock signal is coupled through inverters  90  and  92 , through capacitor-diode AC level shift network  94  and through microstrip resonator L1, to the emitter of microwave oscillator transistor  96 . Whenever the RX clock goes low, transistor  96  is biased on and commences to oscillate within a nanosecond. 
     The same negative going clock edge that biases-on transistor  96  also propagates from inverter  90  through an RC delay network  98  and inverter  100  through another microstrip resonator L1′ to the collector of transistor  96 . Thus, a negative-going edge appears at the collector shortly after the emitter of the oscillator swings negative. The first edge, in the emitter circuit, biases the oscillator on and the second edge, in the collector circuit, biases it off, i.e., drives the collector bias to zero so oscillations cease. The propagation delays of inverters  92  and  100  are matched and cancel, so the difference in edge timing, which sets the oscillation time, is precisely determined by adjustable RC network  98 . Experiments show that the stability of the RF oscillation can be maintained to better than 100 ps in width over a wide temperature range by using this “dual bias-path” approach. This circuit has been described in co-pending application “Pulsed RF Oscillator and Radar Motion Sensor,” Ser. No. 09/073,159, by McEwan. 
     Microwave oscillator transistor  96  is connected to quarter-wave resonator microstrips  102  (of length L1) that define its operating frequency, which is typically 6 GHz or the 4 th  sub-harmonic of 24 GHz. A short microstrip  104  of nonresonant length L3 in the base circuit provides inductance that is converted into a negative resistance by transistor  96 , which is essential for oscillation in this type of oscillator. One of the microstrip resonators has a short wire or metal tab  106  attached to it to fine-tune the oscillator. During production the tab  106  can be bent to vary its capacitance to ground and thereby tune the RF frequency. Alternatively, a metal screw can be brought close to one of the microstrips for fine-tuning. This tuning sets the receiver center frequency to 24.1 GHz, for example. 
     The receiver bandwidth is set by the width of the RF pulse generated by transistor  96 . For a 6 GHz RF pulse width of 2 ns, the receiver bandwidth at the 4 th  harmonic is 500 MHz wide. Wide bandwidth is needed to receive the short pulses employed by the present invention. 
     RF oscillations at 6 GHz are coupled from the oscillator to microstrip  108  of length L1, which is connected along with other microstrips L2 to harmonic sampling diodes  112 , which are connected to receive antenna  116 . Diodes  112  are a series pair for reasons previously described with reference to FIG.  4 . The cathodes of diodes  112  are driven with 6 GHz oscillations from microstrip  108 . The amplitude of the oscillations and the bias conditions on the diodes are such that only the tips of the RF sinusoids drive the diodes into conduction. Typically, the conduction times may be about 20-picoseconds for each of perhaps a dozen sinewave cycles contained in each RF pulse from transistor  96 . Since 20 ps corresponds to a half cycle duration of a 24 GHz RF sinusoid, sampler diodes  112  are able to efficiently sample 24 GHz signals appearing at the sampler input at antenna  116 , i.e., across quarter-wave microstrip  114  of length L2. The microstrips L2 are approximately one-quarter wavelength long at the received signal frequency, i.e., 24 GHz. Antenna  116  is also a resonant antenna, one-quarter wavelength long, at the received signal frequency. 
     The RX clock has a typical frequency of 1-10 MHz, so diodes  112  conduct frequently and cause a steady pulsed current to flow. This current charges capacitor  122  to an equilibrium voltage set by bias resistor  124 . Bias resistor  124  forms a return path for the average rectified detector current; consequently the conduction angle, or fraction of the 6 GHz sinewave tips that drive the diodes into conduction are influenced by the value of the bias resistor  124 . 
     RF signals at 24 GHz that appear across input microstrip  114  add algebraically with the 6 GHz RF pulses from microstrip  108  and cause a modulation in detection voltage across resistor  124 . This detected signal is amplified by amplifier  31  (formed of op amp  126 ) and filtered by bandpass filter  32  to produce a baseband video signal. Microstrip  114  provides input filtering to select the desired harmonic to be sampled, such as 24 GHz. Input filtering is not necessary for operation, but it helps eliminate unwanted out-of-band signals. 
     Quarter wave microstrip  108  provides an RF short at 24 GHz to prevent RF leakage out of the waveguide and to amplifier  31 . It also enhances the detection efficiency of diodes  112 . 
     This circuit uses a silicon bipolar transistor as the local oscillator since silicon bipolar microwave oscillators exhibit much lower noise than GaAsFET microwave oscillators. The noise penalty associated with sub-harmonic sampling is more than compensated-for by the low noise of the silicon bipolar oscillator. Further advantages include low cost, a relaxed printed circuit layout, and freedom from injection locking on external 24 GHz signals. Transistor  96  is a NEC NE68533, Schottky sampling diode  112  is a HP type HSMS-286C, logic inverters  90 , 92 , and  100  are type 74AC04, and the op amp  126  is a TI type TLO71. 
     The operation of this sampler circuit differs from a conventional harmonic sampler in that it has a high impedance output, rather than the usual  50  ohm output. Further, the output circuit is configured to peak detect extremely short RF pulses and hold the peak from one clock cycle to the next in response to controlled timing. This is accomplished by making the capacitance microstrip  108  sufficiently large to not discharge between pulses. In some cases, stray capacitance is sufficient (such as in FIG.  4 ). The sampler integrates detected RF peaks across many RX clock cycles to reduce noise and interference. It is similar in structure to co-pending application “Charge Transfer Wideband Sample-Hold Circuit” Ser. No. 09/084,502, by McEwan. 
     The entire oscillator circuit  26  and sampler  30  are located inside a waveguide beyond cutoff  118  at 6 GHz to reduce spurious radiation at the local oscillator frequency. Thus, waveguide  118  is somewhat higher than one quarter-wavelength high (to allow the quarter-wave antenna to fit) and one-half wavelength wide at 24 GHz. A horn  120  is coupled to the waveguide to provide gain and reduce sidelobe response. In one prototype, the horn flares from 4×6 mm to 20 mm by 50 mm and provides about 18 dB gain at 24 GHz. 
     FIG. 6 a  is a detailed schematic of the pulse driver  18  and the fundamental frequency oscillator  20  of the present invention. A TX clock signal is coupled to an RC differentiation network  130  and then to a fast pulse driver  18 , substantially comprised of driver transistor  132 , which is coupled in a totem pole configuration to 24 GHz oscillator GaAsFET  134 . The rising edge of the TX clock biases-on transistor  132  for a duration set by differentiation network  130 , which is adjustable to produce conduction times of 0 to 10 ns, typically. Accordingly, the RF oscillation time of the GaAsFET can be set from  0  to 10 ns. Quarter-wave microstrip  136  provides a steady +3 to 5 volts DC bias to the GaAsFET. The GaAsFET operates at zero gate-source bias via inductor  138 . When driver transistor  132  switches the GaAsFET source to ground, the GaAsFET automatically becomes biased-on—it is a depletion mode device that fully conducts with zero gate-source bias. Silicon bipolar transistor  132  is a NEC type NE68533 and GaAsFET oscillator transistor  134  is type NE42484A by NEC. 
     A quarter-wave radiator element  144  is connected to the GaAsFET, and both are located inside a waveguide  146  for shielding; waveguide  146  is coupled to a horn  148  to provide gain and reduce sidelobe radiation. The frequency of oscillation is set by a transmission line, or wire,  140  which is somewhat shorter than one-quarter wavelength at 24 GHz and which is connected to the gate and extends toward the drain. The frequency of oscillation is further set in combination with stray coupling capacitance  142 . When element  140  is a wire, it can be bent to adjust stray capacitance  142  and thereby fine-tune the oscillator frequency. 
     FIG. 6 b  provides details of the physical layout of the fundamental frequency oscillator of FIG. 6 a . The layout is based on a 1.78 mm diameter 4-lead ceramic SMT package, such as the NE42484A by Nippon Electric Corp. A hole  164  is drilled in a printed circuit substrate  166  that has metallization  168  on at least one side. The body of the transistor is inserted into the hole up-side-down. The gate lead  152  and source leads  154  are soldered to a common metallized area  150  with a small relief  156  where metallization is backed away to provide the inductance  138  of FIG. 6 a . Resonator wire  140  is soldered to the gate lead  152  and angled to provide a gap with the drain lead  158 , thereby forming a stray coupling capacitance  142 . Bending wire  140  closer to the drain lead lowers the frequency of oscillation. The drain lead  158  is connected to a quarter-wave monopole antenna  144 , which is further located in the waveguide  146  of FIG. 6 a . FIG. 6 b  depicts a unique arrangement to produce near-mm wave oscillations with a SMT transistor having package dimensions on the order of one-quarter wavelength. 
     Although the invention has been described with reference to a 24 GHz system because of the high interest in this frequency, the principles of the invention can be applied to other frequencies, e.g., 10.5 GHz and 38 GHz. The techniques using harmonic transmit pulses and sub-harmonic sampling can be similarly applied. 
     Changes and modifications in the specifically described embodiments can be carried out without departing from the scope of the invention which is intended to be limited only by the scope of the appended claims.