Abstract:
A low cost glass mount vehicle antenna system utilizes, in a preferred embodiment, an over-coupled quasi-TEM mode transmission line coupler having two simple stamped or printed plates stacked over an L shape internal ground plane. The two conductive plates are located on opposite sides of the glass. With proper open circuit terminations on two of the four coupler ports, the backwards and forward coupling signals are redirected/combined and are effectively fed diagonally between the other two ports, thereby achieving through-glass coupling. The over-coupling achieves this efficient through-glass coupling effect at two spaced-apart frequencies (e.g. 800 MHz and 1800 MHz). The coupler is of a low impedance type and can be used with a variety of collinear array and elevated-feed antennas. It features high efficiency, low backwards radiation, and mechanical simplicity.

Description:
RELATED APPLICATION DATA 
     This application is a continuation-in-part of my copending application Ser. No. 08/740,204, filed Oct. 24, 1996, which claims priority from my provisional application Ser. No. 60/008,071, filed Oct. 25, 1995. The disclosures of these prior applications are incorporated by reference. 
    
    
     BACKGROUND OF THE INVENTION 
     The present invention relates generally to the transmission of radio frequency signals through a dielectric wall (e.g. a vehicle window) and is illustrated in the context of a dual-band, glass mount mobile antenna system. 
     Window mounted antennas have been welcome for many years in mobile radio links, especially in 800 MHz cellular telephone service (sometimes known by the acronym “AMPS”) due to their obvious advantages to the consumer. These advantages include the ease of installation and the fact that it is not necessary to drill a hole in the vehicle, which would detract from its value. Others include enhancing the signal strength for better communication quality, and moving radiation outside the vehicle. Much effort has been devoted to designing effective window mounted antenna systems for mobile radio links. 
     A new type of cellular service, known in the United States as PCS, is growing in popularity. This service occupies frequencies between 1500 and 2000 MHz. (In the United States, the PCS band is at 1900 MHz. In Europe this service (termed PCN) is at 1800 MHz. In Japan this service (termed PHS) is at 1500 MHz.). This alternate cellular service creates a potential compatibility problem with the existing, well-established 800 MHz cellular infrastructure. Many effort have been made to address these comparability issues. The most effective solution seems to be the emergence of multi-mode, multi-band handsets that automatically adapt to the service available in a given area. For example, Qualcomm offers a dual-band, dual mode phone known as the QCP-2700, which provides service over both the 800 MHz AMPS band and the 1900 MHz CDMA PCS band. Ericsson has similar offerings, such as its models PD 328 and PD 398, which each provides both AMPS and PCS service. 
     These dual-band handsets pose a significant engineering challenge, namely the design of a single antenna that provides good performance at both the AMPS and PCS bands. This challenge is compounded when the antenna is vehicle-mounted and fed through-glass. The through-glass coupling system must provide high efficiency coupling (and in some instances antenna matching) at both AMPS and PCS frequencies. Moreover, the bandwidth required at each band is large (e.g. up to 11% in the PCS bands), posing a further engineering obstacle. 
     A variety of through-glass feed techniques are known, as illustrated by the cited patents. Many are capacitively-coupled systems. Examples include U.S. Pat. No. 4,089,817 (Kirkendall, 1978), U.S. Pat. No. 4,839,660 (Hadzoglou, 1989), U.S. Pat. No. 4,992,800 (Parfitt), U.S. Pat. No. 4,857,939 (Shimazaki) and U.S. Pat. No. 4,785,305 (Shyu). In addition to capacitive coupling, these systems also generally employ LC impedance matching networks. 
     There are several problems with the foregoing designs. First the capacitive coupling patches cannot be large in comparison with the operating wavelength. Therefore; high impedance coupling (several hundred ohms) cannot be avoided. This leads to high loss due to the leakage of electrical field at high frequencies. Also, at high frequency bands like PCN/PCS, even a small patch no longer behaves as a lumped capacitor element. Due to the thickness of vehicle glass and various stray capacitances, such capacitive coupling circuits can bypass the signal and make it more difficult to match the (typically) high impedance of the antenna to a 50 ohm system. Additionally, the high impedance coupling creates a moisture sensitive structure. U.S. Pat. No. 4,764,773 (Larsen, 1988) describes a better coupling structure to improve performance in the presence of moisture, but it is still subject to the patch size limitation. 
     Design of a vehicle-mounted radiator also poses difficulties at PCS frequencies. Collinear array whips are desirable for mobile service due to their gain in the vertical plane. However, such whips do not have uniform current distribution. The lower section of the array has the highest current and produces the strongest radiation. But in most vehicle mounting arrangements the lower section of the whip is blocked by the vehicle roof, causing severe pattern distortion and deep nulls. This situation becomes worse at the 1.5-2 GHz PCN/PCS bands simply because the length of radiator is only half that at the 800 Mhz hand due to the doubling of the frequency. 
     Elevated-feed whips are sometimes employed to avoid the pattern distortion caused by vehicle roof blockage of radiation. But elevated-feed antennas are not readily matched for broadband operation (i.e. 11% for DCS-1800). Moreover, many such antennas, employing decoupling sleeve or slots, have low impedance feeds (e.g. 50 ohms). High impedance capacitive-feed systems thus pose large impedance transitions. Impedance transformation at PCS frequencies by use of conventional LC circuits is very inefficient due to the high loss of such circuits at these high frequencies. 
     U.S. Pat. No. Re.33,743 (Blaese) proposes a capacitively coupled antenna system for coupling a coaxial cable through glass to a low impedance quarter-wave whip. But in the PCS bands, the suggested antenna is only 1.7″ long. Again, this is completely below the roof line of vehicle, causing severe pattern distortion and deep nulls. 
     To avoid some of the problems associated with capacitive coupling, a coupling arrangement employing resonant cavities has been proposed. U.S. Pat. No. 4,939,484 (Harada), for example, discloses a through-glass coupler employing a pair of tuned helix cavities. Unfortunately, the liarada cavity aperture must be sized to satisfy a ⅓ object frequency criterion, as described in the patent. That is, for 800 MHz, the helix should be designed for 266 MHz. The resulting cavity has a Q of over 1000 and sufficient coupling aperture. But at the 1.8 GHz band, the helix must be designed for 600 MHz. A 600 MHz helix cavity has a small aperture which is nearly half of the cellular band. A significant drop of unloaded Q is unavoidable due to the thin helix, and the coupling coefficient is not sufficient to provide an 11% bandwidth. Other drawbacks of such helix cavity couplers including highly critical tuning characteristics, and difficulties in mass production due to their complex 3D structure. Impedance matching is also difficult to implement in the cavity context. 
     In my U.S. Pat. No. 5,471,222, a pair of TE 01δ  high dielectric, constant-Q Ba-Bd-Ti oxide (ceramic) resonators were employed to overcome various problems of prior art PCS band through-glass couplers. This approach proved to be highly efficient, with insertion losses of only 0.5 dB through 5 mm automobile glass at 1.8 GHz. However, this arrangement proved sensitive to de-tuning in the field. Additionally, it suffered from a high manufacturing cost. 
     In my U.S. Pat. No. 5,451,966, a rectangular slot coupling scheme was employed to replace the expensive Ba-Nd-Ti Oxide ceramic. This arrangement built on the concept of dual-cavity coupling, where coupling is through an aperture. 
     The idea of slot coupling on an MSA (microstrip antenna) originated by Pozar. It provides a means to overcome the narrow band nature generally associated with MSAs. A “doggie bone”-shaped slot suggested by Pozar significantly increases the magnetic polarisability on the slot. This allows a short slot to achieve the necessary coupling while at the same time keeping backward emissions low. 
     Pozar and other researchers&#39; work was basically limited to numerical solutions of the slot-fed microstrip antenna and multilayer arrays on a ground plane. But the bandwidth advantages of this type of MSA can be used to enhance the performance of the planar slot-cavity coupler. 
     In my above-referenced pending application, an annular ring aperture is employed for through-glass coupling. It is understood that in the rectangular slot design, the requirement for a tight coupling coefficient leads to an increase in slot length, which increases the level of backwards radiation. A major advantage of the annular ring aperture coupler over rectangular slot coupling is that it provides an increased coupling coefficient, which is extremely valuable for coupling through a thick dielectric wall. Another advantage is that the relatively radial distribution of E field on an annular ring aperture coupler successfully reduces the so-called “Microstrip Antenna Effect” in the rectangular slot approach. The annular ring is the complementary element to a small loop antenna. It is well known that a small loop antenna has a very low radiation resistance, and thus has a very low radiation efficiency. But the reduction of backwards radiation merited the tradeoff. Feeding was accomplished without any transition by connecting a coplanar waveguide line directly to the center resonant element. 
     More recently, I have improved the annular ring aperture coupler. That design, shown in the attached FIGS.  7 A- 7 D, employs two small circuit boards  201 ,  202 , one of which is single sided. Inside the vehicle, an annular ring  203  is still used, excited by a stub  204 . The coaxial feedline (not shown) connects with its center connector soldered to end  205  of the stub, and its shield soldered to foil  206 . Plated-through holes  210  connect foil  206  to the groundplane  207  on the opposite side of the inside board  201 . On the outside of the vehicle glass, however, the circuit board defines a loaded microstrip  208 , to which the whip antenna attaches at end  209 . The periphery  211  around the microstrip  208  is foil. A matching function is provided by the microstrip; no additional circuitry is required. The outer surface of the outside circuit board has no foil; just a hole through which the whip antenna connects to end  209 . 
     Some of the evolution in recent high-frequency couplers, and their attendant decrease in transmission loss, is shown by the following: 
     For Rectangular slot, the transmission loss are accumulated as: 
     cable---microstrip--- slot---glass---slot---microstrip---i.m.n.---antenna. 
     For annular ring aperture coupler, the transmission loss are accumulated as: 
     cable---microstrip--- annular ring ---glass--- annular ring ---i.m.n.---antenna. 
     For the most recent work on annular ring, the transmission loss are accumulated as: 
     cable---microstrip---annular ring---glass--- loaded microstrip--- antenna.(integrated i.m.n.) 
     Where i.m.n represents impedance matching network. 
     As evidenced by the foregoing, there are numerous approaches for through-glass coupling at high frequencies. However, such approaches uniformly operate over a single, limited frequency band. The aperture coupled designs probably has the widest bandwidth, but even this is much less than one octave. For AMPS/PCS dual-band operation in the United State, the lowest frequency is 824 MHz and the highest is 1990 MHz, yielding a ratio of 2.415. Even in Europe, the ratio is still 2.112. 
     In accordance with a preferred embodiment of the present invention, a nonhomogenous quasi-TEM mode transmission line directional coupler arrangement is adapted to serve as a dual-band through-glass coupler. Such a directional coupler has four ports, but two are left open-circuited. By this arrangement, the signal fed by coaxial cable to one port is re-directed across the coupler to the diagonal port, which connects to the external antenna. The even and odd mode impedances of the coupling device are selected so that an over-coupled 3 dB coupler is realized; the two crossover points are located at the centers of the two frequency bands of interest. 
     This arrangement features very high efficiency since it is a complete distributed design, with no LC circuit elements. Other advantages include its low impedance coupling, and broadband behavior. Moreover, backwards radiation is substantially avoided while maintaining a high coupling coefficient. The coupler is mechanically rugged, easy to manufacture and inexpensive to produce. 
     A dual-resonant whip antenna or coplanar waveguide dipole type antenna is desirably connected to the coupler, thereby achieving a dual-band glass-mounted antenna system. 
     The foregoing and other features and advantages will be more readily apparent from the following detailed description, which proceeds with reference to the accompanying drawings. 
     These objectives are accomplished in the present invention by implementing the quasi-TEM mode transmission line coupler with proper termination, providing an antenna with collinear elements while preserving the performance of the previous arts at the same time. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a perspective illustration of a dual-band antenna assembly according to a preferred embodiment of the present invention. 
     FIG. 2A shows an equivalent circuit for the directional coupler employed in the FIG. 1 embodiment. 
     FIG. 2B is another view of the directional coupler employed in FIG. 1 
     FIG. 2C illustrates the “eye” of the coupling curve that allows two spaced-apart operating bands (or a single operating band, depending on tuning). 
     FIG. 3 is a perspective illustration of a single band antenna system according to a second embodiment of the present invention. 
     FIG. 4 illustrates another embodiment of the present invention wherein an elevated-fed antenna driven by use of a virtual ground plane. 
     FIG.  5   a  is a plot showing the typical VSWR characteristics of the antenna system of FIG.  1 . 
     FIG.  5   b  is a plot showing the typical VSWR characteristics of the antenna system of FIG.  3 . 
     FIGS. 6A and 6B are scale drawings of the first and second plates used in the FIG. 1 embodiment. 
     FIGS.  7 A- 7 D show front and back sides of two circuit boards used in the above-described improved annular ring coupler. 
    
    
     DETAILED DESCRIPTION 
     FIG. 1 shows an exploded view of a dual-band antenna system according to one embodiment of the present invention. 
     Outside assembly  60  has the active whip antenna assembly  80  mounted on it. The housing  60  can be made of some thermal plastic materials such as ABS for rigidity and UV stability. Metal swivel part  67  may be insert-molded inside the housing  60  so that it has robust mechanical strength and is moisture isolated. The swivel member  69  and the whip assembly  80  are fixed onto the housing  60  by screw  70 . This assembly forms a conductive swivel with a locking mechanism so that the angle of the antenna can be adjusted during the initial installation and subsequent re-adjustment in the field. By providing mating thread  69  and  89  on whip assembly  80  and housing  60 , respectively, the whip  80  is detachable for some purpose such as drive-through car wash. The swivel holder member  67  is electrically connected to the edge of outside metal plate member  61  through the extension  62 . 
     A small protrusion  62  from the metal plate member  60  provides an inductive effect which, at 1800 MHz, helps match the capacitive impedance presented by the ⅝ wavelength lower section of the whip  80 . Metal plate member  61  is one of the arm of the directional coupler. 
     Inside assembly  38  comprises a plastic housing  20 , a metal plate  26  with approximately the same form as the counterpart of the outside plate  61 . Metal plate member  26  is the second arm of the directional coupler. An L-shaped metal piece  35  serves as the common ground plane of the coupler (and serves as a shield preventing backwards radiation). The coupler is fed by a coaxial cable  23  with center conductor  25  connected to the inside plate member  26  through extension  31  and the shield connected to the folded up portion  36  of L shape metal  35 . Cable  23  can be any type of popular low loss coaxial cable. The other end of cable  23  is connected to a RF connector  22  which goes to a radio transceiver (not shown). 
     The inside and outside assembly must be aligned for proper operation. There are two alignment conditions, as follows: 
     Conductive plates  26  and  61  (on opposite sides of the vehicle glass) must face each other. The plates may be slightly offset (sideways) from one another. Such offset does not impair the directional coupler mode of operation, and provides and additional degree of freedom in tuning the coupler for best impedance match. 
     The antenna and the feedline must be connected to diagonally opposite ports of the coupler. In the illustrated embodiment, this means that the antenna connects to one edge of one plate, and the feed connects to the remote (opposite) edge of the other plate. In most applications, the inside and outside components of the coupler will be attached by adhesive to opposing sides of the vehicle glass. Once one component is installed, its orientation dictates the orientation at which the other component must be installed. If the inside and outside components are improperly oriented relative to each other, severe mismatch and coupling inefficiency will result. 
     Referring to FIGS. 2A and 2C, plates  61  and  26  cooperate to form a quasi-TEM mode directional coupler. This coupler has four ports, although two are left open-circuited and thus are not obvious from inspection of the physical device. One of the open-circuited ports (port  2  in FIG. 2A) is, in physical terms, the edge of plate  61  opposite protrusion  62 . The other of the open-circuited ports (port  4  in FIG. 2A) is the edge of plate  26  remote from point  31 . (In common parlance, open port  2  is known as the backward coupling port, and open port  4  is known as the through port, although these names are misdescriptive in the present novel use of this coupler.) 
     The feedline is connected to port  1  in FIG. 2A (generally known as the “input port”). This port is the edge of plate  26  to which terminal  31  connects. The antenna is connected to diagonal port  3  (generally known as the “isolation port). This port is the edge of plate  61  from which protrusion  62  extends. 
     The illustrated arrangement of open circuits on ports  2  and  4  causes energy to be diagonally coupled between ports  1  and  3 . 
     The illustrated coupler is nonhomogeneous, resulting in different even and odd mode phase velocities. To increase the directivity of the coupler, a set of small legs or taps  27 ,  28 ,  29 ,  30 ,  63 ,  64 ,  65 ,  66  (FIG. 2B, FIGS. 6) are provided on the edges of the inside and outside plates. 
     The amount of coupling is determined by the distances between the L-shaped metal and the inside plate, the width of the plates, the effective dielectric constant of the window and adhesive assembly and the thickness of the glass and adhesive pads. The operating band is primarily decided by the length of the coupling plate as shown in FIG.  2   b.    
     The inside and outside assemblies are mounted onto the vehicle&#39;s window through adhesive pads sets  41 ,  42  and  43 ,  44 , respectively (FIG.  1 ). Two adhesive patches are employed on each side to permit the planar coupling assemblies to be securely mounted to the (generally) curved vehicle glass. 3M double-sided tape with a thickness of about 1 mm is used in the preferred embodiment. The edges and the open area are desirably sealed by silicone for waterproofing. 
     It is known that stripline broadside-coupled 3 dB directional couplers have good broadband characteristics. However, once transformed to microstrip, one of the ground planes is removed and the TEM mode changes to quasi-TEM mode. Even and odd mode velocities are different due to the different materials and mode change. 
     Referring to FIG. 2C, the dual-band operation of the illustrated coupler is based on the fact that the two crossover points are positioned at the center frequency of the desired bands by manipulation of the coupler&#39;s coupling C (dB). Coupling C is a function of the dimensions of the coupler and adjacent effective dielectric constant. If C (dB) is intentionally increased (i.e. the coupler is over-coupled) the crossover points spread and can be positioned at the centers of two spaced-apart frequency bands. Alternatively, C can be reduced to about 3 dB to yield single band operation (as shown by the dashed/dot line). 
     For a 50 ohm system, the standard microwave circuit design procedures detailed below provide a starting point for the coupler&#39;s parameters. (Ultimately, empirical testing is required to set final dimensions.)          fh   fl     =         1920                 MHz       860                 MHz       =   2.2326               f   0     =         824                 MHz     +     1990                 MHz       2                            
     Any conventional linear simulator can be to get desired coupling value for a strip line model (S. B. Cohn&#39;s original coupler) 
     The next step is to intentionally select an over-coupled C value so that the two crossing points occur at the center frequencies of PCS and AMPS bands, respectively. This can be done graphically. Once C is determined, say −2.5 dB, the coupling coefficient can be obtained. 
     
       
         C(dB)=20log 10 (K) 
       
     
     Then the even and odd mode impedance can be calculated as:          Z     0      e       =       Z   0              1   +   K       1   -   K                     Z     0      o       =       Z   0              1   -   K       1   +   K                                  
     Where Z0 is the impedance of the coaxial cable. Then the dimension can be synthesized as follows:          Z     0      e       =         η   0       2          ɛ   eff                    K   ′          (   k   )         K        (   k   )                     Z     0      o       =     296.1         ɛ   eff              b   sl       S   sl              tanh     -   1            (   k   )                                  
     Where K′(k) is the Elliptical integrals of the 1 st  kind while k is the solution of equation pairs as follows:            W   sl       b   sl       =       1   π          [       1        n        (       1   +   R       1   -   R       )         -         S   sl       b   sl          1        n        (       1   +     R   k         1   -     R   k         )           ]               R   =             kb   sl     /     S   sl       -   1           b   sl     /     (     kS   sl     )       -   1                                
     For each k, the Elliptical Integral can be solved numerically, using computer techniques disclosed, e.g., in Press et al,  Numerical Recipes in C , 2d. ed., Cambridge Univ. Press, 1992. 
     Iteration has to be performed to fit S. Finally the dimensions of microstrip version coupler can be derived:        W   =     W   sl             S   =     S   sl             h   =         b   sl     -     S   sl       4                            
     The illustrated coupler has a non-homogeneous dielectric, including variously air, adhesive, tape, and window. This dielectric is desirably treated as a thick substrate microstrip line where open end effect must be deducted from the length. The length of the coupling fingers can be calculated with end effect taken into account:            λ   g0     /   4     =       C   0       4          ɛ   eff            f   0                   δ                   l        (   x   )         =     (         ξ   1          ξ   3          ξ   5         ξ   4       )               ξ   1     =       0.434907        [         ɛ   eff   0.81     -   0.26         ɛ   eff   0.81     -   0.189       ]            [         x   0.8544     +   0.236         x   0.8544     +   0.87       ]                 ξ   2     =     1   +       x   0.371         2.358        ɛ   eff       +   1                   ξ   3     =     1   +     [       0.5247          tan     -   1            [     0.084        x     1.9413     ɛ      2           ]           ɛ   eff   0.9236       ]                 ξ   4     =     1   +     0.0377            tan     -   1            [     0.067        x   1.456       ]            [     6   -     5        e     0.036        (     1   -     ɛ                 eff       )             ]                     ξ   5     =     1   -     0.218        e       -   7.5        x                                  
     The coupling arm length for outside and inside couplers are expressed as: 
       l   outside =λ g0 /4 −δl ( W /( h+S ))( h+S ) 
     
       
           l   inside =λ g0 /4 −δl ( W/h ) h   
       
     
     Since no significant difference is observed, the same length is used for the inside and outside coupling plates. 
     After initial data is calculated, a full-wave numerical simulation can be performed to tweak and optimize the performance since, in reality, the situation is much more complex than the idealized situation modeled by these equations. A Integration Equation Method based MoM (Method of Moment) 3D RF/Microwave structure simulation software IE3D™ (Zeland Software Inc., Fremont, Calif.) is a preferred simulation tool. It has been found that with the folding of the ground plate for adaptation of the coaxial cable, the electrical length of the coupling plates must be reduced to compensate the center frequency shift. 
     In the preferred embodiment, each coupling plate measures 22 mm by 24 mm, exclusive of the taps. (Suitable performance can be achieved without the taps, particularly at higher frequencies.) The ground plate  35  measures 40 mm (Wg) by 45 mm (Lg), with the cable side folded to form an L-shape so that a coaxial to microstrip transition can be made. The folded-up portion is about 12.5 mm. The spacing between the ground plate  35  and the inside coupling plate is also about 12.55 mm. 
     In the preferred embodiment, a stub  99  extends from the end of plate  35  opposite the fold to balance the ground current for the lower band since there is no ground plane for the on glass antenna. A 55 mm wire having 1 mm diameter is used in the preferred embodiment. 
     Back to FIG. 1, the whip assembly  80  is a collinear array type with a single-feeding point provided by coupler output  62 . Assembly  80  includes with top radiator elements  85  and  83 . Element  83  is a reverse choke which works together with radiator  85  to form a sleeve type of antenna section. Element  83  can be a standard metal tube and measures about ⅝ wavelength for the higher band and has a diameter of about 8.7 mm. Element  83  is open end at the top but is shorted with whip  85  at the bottom. A cylindrical lower radiator member  81  and the swivel members  69 ,  67  form the lower section of the whip assembly. The two radiators are separated by an air-wounded phasing coil  82 . Desirably, coil  82  and whip  85  are formed from a unitary piece of metal (e.g. copper or stainless steel) having a diameter of about 1.8 mm. The whole whip assembly is encapsulated with low loss plastic material, either by a plastic shell or completely molded together. 
     For the higher frequency band, the radiator member  85  and  83  provide in-phase radiation. The lower section has the same phase as the upper section by means of the phasing coil  82 . Therefore at least 2.5 in-phase dipoles are furnished for the higher band. The feed impedance on the higher band is close to a ⅝ radiator due to the current distribution. This capacitive reactance is countered by the inductance provided by protrusion  62  from plate  61 , as mentioned earlier. 
     For the lower frequency band, the inductance of stub  62  is negligible. Upper radiators  85  and  83  are still in phase since these elements cooperate to define a reverse choke. The phase starts reversing at the upper to middle section of the lower radiator  81  and increases along the bottom, making it a “current-fed antenna” with an impedance of about 50 ohms. Considering the proportion of the current distribution, it still has strong low angle radiation but the pattern splits at about 15 degrees of elevation angle. 
     Whip  80  thus provides a collinear dual band array that is current-fed at the lower frequency band and voltage fed at the higher frequency band, thereby facilitating relative independent tuning. 
     As stated earlier, the coupler employed in FIG. 1 can be designed to provide single band operation, if desired. Such a coupler is advantageous due to its simplicity and efficiency, whether at 800 MHz, 1800 MHz, or elsewhere. 
     The coupling factor C in dB for a single-band operation is selected either the way that maximum coupling occurs at the center of desired band or the way in a dual-band design described previously. There is always a trade-off between size and which crossover portion being used. For example, C=−2.9 dB results in a more than 10% bandwidth. 
     FIG. 3 shows the detailed construction of the single band PCS/PCN antenna system. The coupler plates are the same for higher frequency band since the 2 nd  crossover is available. The whip assembly  160  can be a ½ wavelength whip section  162  stacked over a ⅝ wavelength section ( 165 ,  166 ,  142 ,  143 ) through a 180 degree phasing coil  164 . Again the coil is encapsulated for environmental reasons. The extension  146  of the outside plate  145  serves as an inductor for the matching of ⅝ wave base section. Notches  147  and  148  on the outside plate  145  can effectively reduce the size of the inductance  146  trace length. 
     One advantage of the illustrated coupler is that a virtual ground plane can be provided outside the vehicle glass. This facilitates use with elevated-feed antennas, such as sleeve dipoles. 
     Referring to FIG. 4, the ground plate  35  is extended lengthwise to underlie another metal plate  180  outside the window. Edges  185  and  186  are aligned together. The additional patch  180  is placed aside the main coupling plate  61 . The outside plate  180  and the inside extended plane  35  are separated by the window. If a ¼ wavelength dimension is selected for the outside plate  180  and the ends are open, edge  182  of outside plate  180  is the short circuit due to the quarter-wave transformation. Therefore a “virtual ground point” is realized at this point. An elevated-fed antenna can be fed between this edge  182  and the coupler edge  181 . At least one band can be covered for high feeding point or a compromised performance for dual-band operation. 
     FIG.  5   a  shows the typical VSWR of the dual-band antenna system of FIG.  1 . FIG.  5   b  is a similar plot but for the single band antenna system of FIG.  3 . 
     While the foregoing discussion has described the conductive plates as being metal sheets, in other embodiment circuit board implementations can naturally be used. Likewise, while the whip antenna has been shown as being wire based, the whip, too, can be fabricated as a blade using a planar etched printed circuit. 
     Having described and illustrated the principles of my invention with reference to a preferred embodiment, and various alternatives thereof, it should be apparent that my invention can be modified in arrangement and details without departing from such principles.