Abstract:
A digital coherent receiving apparatus includes a first oscillator for outputting a local light signal of a fixed frequency, a hybrid unit mixing the local light signal with a light signal received by a receiver, a second oscillator for outputting a sampling signal of a sampling frequency, a converter for converting the mixed light signal into digital signal synchronizing with the sampling signal, a waveform adjuster for adjusting a waveform distortion of the converted digital signal, a phase adjustor for adjusting a phase of the digital signal adjusted by the waveform adjustor, a demodulator for demodulating the digital signal adjusted by the phase adjuster, and a phase detector for detecting a phase of the digital signal adjusted by the phase adjuster, and a control signal output unit for outputting a frequency control signal on the basis of the detected phase signal to the second oscillator.

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
       [0001]    This application is based upon and claims the benefit of priority of the prior Japanese Patent Application No. 2009-150124, filed on Jun. 24, 2009, the entire contents of which are incorporated herein by reference. 
       FIELD 
       [0002]    The embodiments discussed herein are a digital coherent receiver. 
       BACKGROUND 
       [0003]    Along with an increase in the internet traffic, a larger capacity in an optical communication system of a trunk line system is demanded, and research and development are carried out on an optical transmitter receiver capable of transmitting a signal exceeding 100 [Gbit/s] per wavelength. However, in the optical communication, when a bit rate per wavelength is increased, a degradation in a signal quality becomes large because of an decrease in the bearing force for Optical Signal Noise Ratio (OSNR), the wavelength dispersion in a transmission path, the polarized wave mode dispersion, or the waveform distortion from an nonlinear effect or the like. 
         [0004]    For this reason, in recent years, a digital coherent reception system having the OSNR bearing force and the waveform bearing force in the transmission path attracts attention (for example, see D. Ly-Gagnon et al, IEEE JLT, vol. 24, pp. 12-21, 2006). Also, in contrast to a system for the direct detection by assigning ON/OFF of a light intensity to binary signals in a related art, according to the digital coherent reception system, a light intensity and phase information are extracted through the coherent reception system. Then, the extracted intensity and phase information are quantized by an ADC (Analog/Digital Converter) and demodulated by a digital signal processing circuit (for example, see F. M. Gardner, “A BPSK/QPSK timing-error detector for sampled receivers”, IEEE Trans. Commun., vol. COM-34, pp. 423-429, May 1986). 
         [0005]    However, according to the related art technology, when the frequency of the local light in the digital coherent receiver varies with respect to the frequency of the optical light transmitted from the transmitter, the optical signal cannot be digitally demodulated at a satisfactory precision in the digital coherent receiver. For this reason, a problem occurs that a communication quality is degraded. 
       SUMMARY 
       [0006]    According to an aspect of the invention, A digital coherent receiving apparatus includes a receiver for receiving a light signal, a first oscillator for outputting a local light signal of a fixed frequency, a hybrid unit mixing the local light signal output from the first oscillator with the light signal received by the receiver, a second oscillator for outputting a sampling signal of a sampling frequency, the second oscillator changing the sampling frequency in response to an input a frequency control signal, a converter for converting the mixed light signal into digital signal by sampling the mixed light signal synchronizing with the sampling signal, a waveform adjuster for adjusting a waveform distortion of the digital signal converted by the convertor, a phase adjustor for adjusting a phase of the digital signal adjusted by the waveform adjustor, a demodulator for demodulating the digital signal adjusted by the phase adjuster, a phase detector for detecting a phase of the digital signal adjusted by the phase adjuster; and a control signal output unit for outputting a frequency control signal on the basis of the detected phase signal to the second oscillator. 
         [0007]    The object and advantages of the invention will be realized and attained by means of the elements and combinations particularly pointed out in the claims. 
         [0008]    It is to be understood that both the foregoing general description and the following detailed description are exemplary and explanatory and are not restrictive of the invention, as claimed. 
     
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         [0009]      FIG. 1  is a block diagram showing a configuration example 1 of a digital coherent receiver; 
           [0010]      FIG. 2  is a block diagram showing a configuration example 2 of the digital coherent receiver; 
           [0011]      FIG. 3  is a block diagram showing a configuration example 3 of the digital coherent receiver; 
           [0012]      FIG. 4  is a block diagram showing a configuration example 4 of the digital coherent receiver; 
           [0013]      FIG. 5  is a block diagram showing a specific example 1 of the phase control circuit shown in  FIGS. 1 to 3 ; 
           [0014]      FIG. 6  is a block diagram showing a specific example 2 of the phase control circuit shown in  FIGS. 1 to 3 ; 
           [0015]      FIG. 7  is a block diagram showing a specific example 1 of a phase adjuster; 
           [0016]      FIG. 8  is a block diagram showing a specific example 2 of the phase adjuster; 
           [0017]      FIG. 9  is a block diagram showing a specific example of a first DLF; 
           [0018]      FIG. 10  is a block diagram showing a specific example of a second DLF; 
           [0019]      FIG. 11  is a block diagram showing a specific example 3 of the phase control circuit shown in  FIGS. 1 to 3 ; 
           [0020]      FIG. 12  is a block diagram showing a specific example 4 of the phase control circuit shown in  FIGS. 1 to 3 ; 
           [0021]      FIG. 13  is a block diagram showing a specific example 1 of a compensation circuit; 
           [0022]      FIG. 14  is a block diagram showing a specific example 2 of the compensation circuit; 
           [0023]      FIG. 15  is a block diagram showing a specific example 1 of the phase control circuit shown in  FIG. 4 ; 
           [0024]      FIG. 16  is a block diagram showing a specific example 2 of the phase control circuit shown in  FIG. 4 ; 
           [0025]      FIG. 17  is a block diagram showing a specific example 3 of the phase control circuit shown in  FIG. 4 ; 
           [0026]      FIG. 18  is a block diagram showing a specific example 4 of the phase control circuit shown in  FIG. 4 ; 
           [0027]      FIG. 19  is a block diagram showing a specific example 1 of a frequency/phase compensation circuit; 
           [0028]      FIG. 20  is a block diagram showing a specific example 2 of the frequency/phase compensation circuit; 
           [0029]      FIG. 21  is a block diagram showing a specific example 5 of the phase control circuit shown in  FIGS. 1 to 3 ; 
           [0030]      FIG. 22  is a block diagram showing a specific example 6 of the phase control circuit shown in  FIGS. 1 to 3 ; 
           [0031]      FIG. 23  is a block diagram showing a configuration example of a phase detector used for a phase detection unit; 
           [0032]      FIG. 24  is a graph showing a sensitivity correction by a phase detector of a sensitivity correction type (single-sided correction); 
           [0033]      FIG. 25  is a graph showing a sensitivity correction by a phase detector of a sensitivity correction type (two-sided correction); 
           [0034]      FIG. 26  is a block diagram showing a configuration example of a sensitivity monitor phase detector (single-sided monitor); 
           [0035]      FIG. 27  is a block diagram showing a configuration example of a sensitivity monitor phase detector (two-sided monitor); 
           [0036]      FIG. 28  is a block diagram showing a configuration example of a phase detection unit of a sensitivity selection correction type; 
           [0037]      FIG. 29  is a block diagram showing a configuration example 1 of a phase detection unit of a diversity addition type; 
           [0038]      FIG. 30  is a block diagram showing a configuration example 2 of the phase detection unit of the diversity addition type; 
           [0039]      FIG. 31  is a block diagram showing a configuration example 3 of the phase detection unit of the diversity addition type; 
           [0040]      FIG. 32  is a block diagram showing a configuration example 4 of the phase detection unit of the diversity addition type; 
           [0041]      FIG. 33  is a block diagram showing a specific example of an equalization filter (polarized wave dispersion equalization); 
           [0042]      FIG. 34  is a block diagram showing a specific example of an equalization filter (wavelength dispersion equalization); 
           [0043]      FIG. 35  is a block diagram showing a modified example 1 of the digital coherent receiver; 
           [0044]      FIG. 36  is a block diagram showing a modified example 2 of the digital coherent receiver; 
           [0045]      FIG. 37  is a block diagram showing a specific example of a frequency difference detector; 
           [0046]      FIG. 38  is a block diagram showing a specific example of a frequency compensator; 
           [0047]      FIG. 39  is a block diagram showing a specific example of an optical transmission system; 
           [0048]      FIG. 40  is a block diagram showing specific example of a Fourier transform unit and an inverse Fourier transform unit; and 
           [0049]      FIG. 41  shows an operation of the circuit shown in  FIG. 40 . 
       
    
    
     DESCRIPTION OF EMBODIMENTS 
       [0050]    Hereinafter, with reference to the attached drawings, preferred embodiments of this digital coherent receiver will be described in detail. 
         [0051]    (Degradation in Communication Quality Due to Frequency Fluctuation) 
         [0052]    First, degradation in a communication quality due to a frequency fluctuation of a local light source will be described. In a configuration in which the wavelength dispersion compensation is carried out by a waveform distortion compensator of the digital coherent receiver, in proportion to the size of the wavelength dispersion amount to be compensated in the waveform distortion compensator, a phenomenon is generated that the frequency fluctuation of the local light source is transformed into the sampling phase fluctuation. 
         [0053]    A specific description will be given of this phenomenon. The transmission signal transmitted from the optical transmitter can be represented, for example, by the following expression (1). In the following expression (1), s(t) denotes a modulation signal for generating the transmission signal. Denoted by j is an imaginary number. Denoted by t is time. Denoted by ω o  is a carrier wave frequency of the light. 
         [0000]      [Expression 1] 
         [0000]      s(t)exp(jω 0 t)  (1)
 
         [0054]    A transfer function of the transmission path dispersion can be represented, for example, by the following expression (2). In the following expression (2), D denotes wavelength dispersion. V L  denotes a light speed. Denoted by ω are respective frequencies of the baseband. 
         [0000]    
       
         
           
             
               
                 
                   [ 
                   
                     Expression 
                      
                     
                         
                     
                      
                     2 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   exp 
                   [ 
                   
                     j 
                      
                     
                         
                     
                      
                     
                       
                         π 
                          
                         
                             
                         
                          
                         
                           V 
                           L 
                         
                          
                         D 
                       
                       
                         ω 
                         0 
                         2 
                       
                     
                      
                     
                       
                         ( 
                         
                           ω 
                           - 
                           
                             ω 
                             0 
                           
                         
                         ) 
                       
                       2 
                     
                   
                   ] 
                 
               
               
                 
                   ( 
                   2 
                   ) 
                 
               
             
           
         
       
     
         [0055]    The reception signal distorted by the wavelength dispersion can be represented by the following expression (3). 
         [0000]    
       
         
           
             
               
                 
                   [ 
                   
                     Expression 
                      
                     
                         
                     
                      
                     3 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       
                           
                         ⋀ 
                       
                        
                       S 
                     
                      
                     
                       ( 
                       
                         ω 
                         - 
                         
                           ω 
                           0 
                         
                       
                       ) 
                     
                   
                    
                   
                     exp 
                     [ 
                     
                       j 
                        
                       
                           
                       
                        
                       
                         
                           
                             πV 
                             L 
                           
                            
                           D 
                         
                         
                           ω 
                           0 
                           2 
                         
                       
                        
                       
                         
                           ( 
                           
                             ω 
                             - 
                             
                               ω 
                               0 
                             
                           
                           ) 
                         
                         2 
                       
                     
                     ] 
                   
                 
               
               
                 
                   ( 
                   3 
                   ) 
                 
               
             
           
         
       
     
         [0056]    ̂S denotes a frequency domain display of the transmission modulation signal. The local light can be represented by the following expression (4). In the following expression (4), Δω denotes a frequency difference between the signal light and the local light. 
         [0000]      [Expression 4] 
         [0000]      exp[ j (ω 0 −Δω) t]   (4)
 
         [0057]    The signal after the coherent reception after the local light and the signal light shown in the expression (4) are mixed can be represented as the following expression (5). 
         [0000]    
       
         
           
             
               
                 
                   [ 
                   
                     Expression 
                      
                     
                         
                     
                      
                     5 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       
                           
                         ⋀ 
                       
                        
                       S 
                     
                      
                     
                       ( 
                       
                         ω 
                         - 
                         Δω 
                       
                       ) 
                     
                   
                    
                   
                     exp 
                     [ 
                     
                       j 
                        
                       
                           
                       
                        
                       
                         
                           
                             πV 
                             L 
                           
                            
                           D 
                         
                         
                           ω 
                           0 
                           2 
                         
                       
                        
                       
                         
                           ( 
                           
                             ω 
                             - 
                             Δω 
                           
                           ) 
                         
                         2 
                       
                     
                     ] 
                   
                 
               
               
                 
                   ( 
                   5 
                   ) 
                 
               
             
           
         
       
     
         [0058]    In the digital coherent receiver, the signal represented by the expression (5) is quantized by the ADC to carry out the digital signal processing. The following expression (6) represents an inverse transfer function of the transmission path dispersion in a case where the dispersion compensation in the waveform compensation circuit of the digital signal processing circuit is performed. In the following expression (6), AD denotes a deviation of the transmission path dispersion and the dispersion compensation amount compensated in the waveform compensation circuit. 
         [0000]    
       
         
           
             
               
                 
                   [ 
                   
                     Expression 
                      
                     
                         
                     
                      
                     6 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   exp 
                   [ 
                   
                     
                       - 
                       j 
                     
                      
                     
                       
                         
                             
                         
                          
                         
                           
                             πV 
                             L 
                           
                            
                           
                             ( 
                             
                               D 
                               - 
                               
                                 Δ 
                                  
                                 
                                     
                                 
                                  
                                 D 
                               
                             
                             ) 
                           
                         
                       
                       
                         ω 
                         0 
                         2 
                       
                     
                      
                     
                       ω 
                       2 
                     
                   
                   ] 
                 
               
               
                 
                   ( 
                   6 
                   ) 
                 
               
             
           
         
       
     
         [0059]    The following expression (7) represents the signal after the wavelength dispersion compensation. 
         [0000]    
       
         
           
             
               
                 
                   [ 
                   
                     Expression 
                      
                     
                         
                     
                      
                     7 
                   
                   ] 
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     
                       
                           
                         ⋀ 
                       
                        
                       S 
                     
                      
                     
                       ( 
                       
                         ω 
                         - 
                         Δω 
                       
                       ) 
                     
                   
                    
                   
                     exp 
                     [ 
                     
                       j 
                        
                       
                           
                       
                        
                       
                         
                           πV 
                           L 
                         
                         
                           ω 
                           0 
                           2 
                         
                       
                        
                       
                         ( 
                         
                           
                             Δ 
                              
                             
                                 
                             
                              
                             D 
                              
                             
                                 
                             
                              
                             
                               ω 
                               2 
                             
                           
                           - 
                           
                             2 
                              
                             D 
                              
                             
                                 
                             
                              
                             
                               Δω 
                               · 
                               ω 
                             
                           
                           + 
                           
                             D 
                              
                             
                                 
                             
                              
                             Δ 
                              
                             
                                 
                             
                              
                             
                               ω 
                               2 
                             
                           
                         
                         ) 
                       
                     
                     ] 
                   
                 
               
               
                 
                   ( 
                   7 
                   ) 
                 
               
             
           
         
       
     
         [0060]    In the expression (7), when a consideration is given on ΔD=0, the signal after the waveform compensation can be represented by the following expression (8). 
         [0000]    
       
         
           
             
               
                 
                   
                       
                   
                    
                   
                     [ 
                     
                       Expression 
                        
                       
                           
                       
                        
                       8 
                     
                     ] 
                   
                 
               
               
                 
                     
                 
               
             
             
               
                 
                   
                     s 
                     ( 
                     
                       t 
                       - 
                       
                           
                       
                        
                       
                         
                           2 
                            
                           
                             
                               πV 
                               L 
                             
                              
                             
                               ( 
                               
                                 D 
                                  
                                 
                                     
                                 
                                  
                                 Δ 
                                  
                                 
                                     
                                 
                                  
                                 ω 
                               
                               ) 
                             
                           
                         
                         
                           ω 
                           0 
                           2 
                         
                       
                     
                     ) 
                   
                    
                   
                     exp 
                     [ 
                     
                       jΔω 
                       ( 
                       
                         t 
                         - 
                         
                             
                         
                          
                         
                           
                             2 
                              
                             
                               
                                 πV 
                                 L 
                               
                                
                               
                                 ( 
                                 
                                   D 
                                    
                                   
                                       
                                   
                                    
                                   Δ 
                                    
                                   
                                       
                                   
                                    
                                   ω 
                                 
                                 ) 
                               
                             
                           
                           
                             ω 
                             0 
                             2 
                           
                         
                       
                       ) 
                     
                     ] 
                   
                    
                   
                     exp 
                     ( 
                     
                       j 
                        
                       
                         
                           π 
                            
                           
                               
                           
                            
                           
                             
                               V 
                               L 
                             
                              
                             
                               ( 
                               
                                 D 
                                  
                                 
                                     
                                 
                                  
                                 Δ 
                                  
                                 
                                     
                                 
                                  
                                 
                                   ω 
                                   2 
                                 
                               
                               ) 
                             
                           
                         
                         
                           ω 
                           0 
                           2 
                         
                       
                     
                     ) 
                   
                 
               
               
                 
                   ( 
                   8 
                   ) 
                 
               
             
           
         
       
     
         [0061]    From the expression (8), because of the frequency drift of the signal light and the local light and the wavelength dispersion compensation, it is understood that a delay of 2πV L DΔω/ω 0   2  is generated. In this manner, in proportion to the size of the wavelength dispersion amount to be compensated in the waveform distortion compensator, the frequency fluctuation of the local light source is transformed into the sampling phase fluctuation, which affects the accuracy of the digital demodulation in the subsequent stage. 
         [0062]    (Influence on ADC Sampling Frequency Control Processing) 
         [0063]    Also, in the digital coherent reception system where the bit rate is equal to or larger than several tens of Gbit/s, the ADC sampling frequency also becomes equal to or larger than several tens of GHz. For this reason, in a case where a digital signal processing circuit is constructed by using an inexpensive CMOS (Complementary Metal Oxide Semiconductor) process, a serial parallel conversion of the ADC sampling signal is performed so that the operation frequency becomes about several hundreds of MHz, and the digital signal processing is performed on the paralleled reception signals. In this manner, in a case where the high speed sampling is carried out in the ADC, the circuit scale becomes larger. 
       EMBODIMENTS 
       [0064]      FIG. 1  is a block diagram showing a configuration example 1 of a digital coherent receiver. As shown in  FIG. 1 , a digital coherent receiver  100  according to an embodiment is provided with a PBS  111 , a local light source  112 , a PBS  113 , a hybrid circuit  121 , a hybrid circuit  122 , photoelectric converters  131  to  134 , a frequency variable oscillator  140 , a digital conversion unit  150 , and a digital signal processing circuit  160 . The digital coherent receiver  100  is a digital coherent receiver configured to convert a detection result of a signal light from an optical transmission path and a local light into a digital signal to conduct a digital processing. 
         [0065]    To the PBS  111  (Polarization Beam Splitter), the signal light sent via the optical transmission path is input. The PBS  111  separates the input signal light into respective polarization axes (which are set as H axis and V axis). The PBS  111  outputs the separated signal light on the H axis (horizontally polarized wave) to the hybrid circuit  121 . Also, the PBS  111  outputs the separated signal light on the V axis (vertically polarized wave) to the hybrid circuit  122 . 
         [0066]    The local light source  112  generates the local light to be output to the PBS  113 . The PBS  113  separates the local light output from the local light source  112  into respective the respective polarization axes (which are set as H axis and V axis). The PBS  113  outputs the separated local light on the H axis to the hybrid circuit  121 . Also, the PBS  113  outputs the separated local light on the V axis to the hybrid circuit  122 . 
         [0067]    The hybrid circuit  121  (90° optical hybrid) performs detection on the basis of the signal light on the H axis output from the PBS  111  and the local light output from the PBS  113 . The hybrid circuit  121  outputs an optical signal corresponding to an amplitude and a phase in an I channel of the signal light to the photoelectric converter  131 . Also, the hybrid circuit  121  outputs an optical signal corresponding to an amplitude and a phase in a Q channel of the signal light to the photoelectric converter  132 . 
         [0068]    The hybrid circuit  122  (90° optical hybrid) performs detection on the basis of the signal light on the V axis output from the PBS  111  and the local light output from the PBS  113 . The hybrid circuit  122  outputs an optical signal corresponding to the amplitude and the phase in the I channel of the signal light to the photoelectric converter  133 . Also, the hybrid circuit  122  outputs an optical signal corresponding to the amplitude and the phase in the Q channel of the signal light to the photoelectric converter  134 . 
         [0069]    Each of the photoelectric converter  131  and the photoelectric converter  132  photo electrically converts the light signal output from the hybrid circuit  121  be output to the digital conversion unit  150 . Each of the photoelectric converter  133  and the photoelectric converter  134  photo electrically converts the light signal output from the hybrid circuit  122  to be output to the digital conversion unit  150 . 
         [0070]    The frequency variable oscillator  140  (oscillation unit) generates a variable frequency clock to be output to the digital conversion unit  150 . Also, the frequency variable oscillator  140  changes the frequency of the generated clock on the basis of a control of the digital signal processing circuit  160 . 
         [0071]    The digital conversion unit  150  is provided with ADCs  151  to  154 . The ADC  151  digitally samples the signal output from the photoelectric converter  131 . Similarly, the ADCs  152  to  154  respectively digitally samples the signals output from the photoelectric converters  132  to  134 . Also, each of the ADCs  151  to  154  performs the digital sampling in synchronization with the clock output from the frequency variable oscillator  140 . Each of the ADCs  151  to  154  outputs the digitally sampled signal to the digital signal processing circuit  160 . 
         [0072]    The digital signal processing circuit  160  is provided with the waveform distortion compensation circuit  161  (waveform distortion compensation unit), a phase control circuit  162 , and an adaptive equalization type demodulation circuit  163  (demodulation unit). The waveform distortion compensation circuit  161 , the phase control circuit  162 , and the adaptive equalization type demodulation circuit  163  may be realized by one DSP (Digital Signal Processor) or mutually different DSPs. 
         [0073]    The waveform distortion compensation circuit  161  compensates a waveform distortion of the signals output from the ADCs  151  to  154  (distortion generated in the optical transmission path). To be more specific, in the waveform distortion compensation circuit  161 , a semi-static transmission path waveform distortion component which changes depending on a propagation characteristic fluctuation such as temperature fluctuation is compensated. The waveform distortion compensation circuit  161  outputs the respective signal in which the waveform distortion is compensated to the phase control circuit  162 . the waveform distortion compensation circuit  161  may be constructed by one circuit block or may have a cascade connection configuration with a plurality of divided waveform distortion compensation circuit blocks. 
         [0074]    The phase control circuit  162  conducts a digital phase compensation on the respective signals output from the waveform distortion compensation circuit  161 . The phase control circuit  162  outputs the compensated respective signals to the adaptive equalization type demodulation circuit  163 . the phase control circuit  162  may be constructed by one circuit for processing the respective signals from the waveform distortion compensation circuit  161  in parallel or may be constructed by a plurality of circuits corresponding to the respective signals from the waveform distortion compensation circuit  161 . Also, on the basis of the phases of the respective signals output from the waveform distortion compensation circuit  161 , the phase control circuit  162  controls the frequency of the clock output by the frequency variable oscillator  140 . 
         [0075]    The adaptive equalization type demodulation circuit  163  conducts the demodulation on the respective signals output from the phase control circuit  162 . Also, the adaptive equalization type demodulation circuit  163  performs an adaptive equalization type waveform distortion compensation on the respective signals output from the phase control circuit  162  before the demodulation. To be more specific, the adaptive equalization type demodulation circuit  163  compensates the waveform distortion component which is generated in the transmission path and fluctuates at a high speed. the adaptive equalization type demodulation circuit  163  may be constructed by one circuit block or may have a cascade connection configuration with a plurality of adaptive equalization circuit blocks. 
         [0076]    For example, in a case where the ADCs  151  to  154  conduct the digital sampling at equal to or higher than several tens of GHz, such a configuration may be adopted that a multiple PLL (Phase-Locked Loop) using the clock output from the frequency variable oscillator  140  as the reference is provided. Also, the digital coherent receiver  100  shown in  FIG. 1  can cope with both a polarized wave multiplex transmission system multiplexing transmission signals for every polarized wave axis and a non-polarized wave multiplex transmission with which the polarized wave multiplexing of the transmission signals is not carried out. 
         [0077]    In this manner, as the digital coherent receiver  100  detects the phase of the signal in the subsequent stage of the waveform distortion compensation circuit  161 , the phase fluctuation generated in the waveform distortion compensation circuit  161  due to the frequency fluctuation of the local light source  112  can be detected. Also, by compensating the detected frequency fluctuation in the former stage of the adaptive equalization type demodulation circuit  163  to precisely conduct the digital demodulation in the adaptive equalization type demodulation circuit  163 , it is possible to improve the communication quality. 
         [0078]    Also, on the basis of the detected phase of the signal in the subsequent stage of the waveform distortion compensation circuit  161 , the digital coherent receiver  100  controls the sampling phase in the digital conversion unit  150 . To be more specific, the digital coherent receiver  100  controls the frequency of the clock oscillated by the frequency variable oscillator  140 . With this configuration, while the enlargement in the circuit scale is suppressed, it is possible to carry out the high speed sampling in the digital conversion unit  150 . Also, the deviation and wander of the modulation frequency of the optical signal and the sampling frequency in the digital coherent receiver  100  is compensated, and it is possible to reduce the phase compensation amount in the waveform distortion compensation circuit  161 . 
         [0079]    Also, the adaptive equalization type demodulation circuit  163  of the digital coherent receiver  100  compensates the waveform distortion fluctuating at a speed higher than the waveform distortion compensated in the waveform distortion compensation circuit  161  to carry out the demodulation. For example, the waveform distortion compensation circuit  161  compensates the waveform distortion of a semi-static characteristic which changes under the temperature fluctuation or the like. With this configuration, while compensating the phase fluctuation due to the frequency of the transmission light source and the frequency drift of the local light source  112  generated under the temperature fluctuation or the like is compensated in the waveform distortion compensation circuit  161 , it is possible to carry out the high precision waveform distortion compensation and demodulation in the adaptive equalization type demodulation circuit  163 . 
         [0080]      FIG. 2  is a block diagram showing a configuration example 2 of the digital coherent receiver. In  FIG. 2 , a part similar to the configuration shown in  FIG. 1  is assigned with the similar reference symbol, and a description thereof is omitted. As shown in  FIG. 2 , the digital coherent receiver  100  may be provided with a fixed-frequency oscillator  211  and a DDS  212  (Direct Digital Synthesizer) instead of the frequency variable oscillator  140  shown in  FIG. 1 . 
         [0081]    The fixed-frequency oscillator  211  (oscillation unit) generates a fixed frequency clock to be output to the DDS  212 . On the basis of the clock output from the fixed-frequency oscillator  211 , the DDS  212  generates a clock to be supplied to the digital conversion unit  150  as a sampling control clock. Also, the DDS  212  changes the frequency of the clock to be generated on the basis of the control of the digital signal processing circuit  160 . Each of the ADCs  151  to  154  performs the digital sampling in synchronism with the clock output from the DDS  212 . 
         [0082]    In this manner, the digital coherent receiver  100  controls the frequency of the sampling control clock supplied by the DDS. With this configuration, while the enlargement in the circuit scale is suppressed, it is possible to carry out the high speed sampling in the digital conversion unit  150 . 
         [0083]      FIG. 3  is a block diagram showing a configuration example 3 of the digital coherent receiver. In  FIG. 3 , a part similar to the configuration shown in  FIG. 1  is assigned with the similar reference symbol, and a description thereof is omitted. As shown in  FIG. 3 , the digital coherent receiver  100  in the case of the non-polarized wave multiplex system may have a configuration provided with a polarized wave controller  311  instead of the PBS  111 , the PBS  113 , the hybrid circuit  122 , the photoelectric converters  133  and  134  and the ADCs  153  and  154  shown in  FIG. 1 . 
         [0084]    The local light source  112  outputs the generated local light to the polarized wave controller  311 . The polarized wave controller  311  controls the polarized wave of the local light output from the local light source  112  so as to be a polarized wave of the signal light received by the digital coherent receiver  100  (for example, H axis). The polarized wave controller  311  outputs the local light in which the polarized wave is controlled to the hybrid circuit  121 . To the hybrid circuit  121 , the signal light sent via the optical transmission path and the local light output from the polarized wave controller  311  are input. such a configuration may be adopted that instead of the frequency variable oscillator  140  shown in  FIG. 3 , the fixed-frequency oscillator  211  and the DDS  212  (see  FIG. 2 ) is provided. the polarized wave controller  311  may be applied to the signal light sent via the optical transmission path instead of the local light. 
         [0085]      FIG. 4  is a block diagram showing a configuration example 4 of the digital coherent receiver. In  FIG. 4 , a part similar to the configuration shown in  FIG. 1  is assigned with the similar reference symbol, and a description thereof is omitted. As shown in  FIG. 4 , the digital coherent receiver  100  may be provided with a fixed-frequency oscillator  411  and a frequency/phase compensation circuit  412  instead of the frequency variable oscillator  140 . 
         [0086]    The fixed-frequency oscillator  411  outputs the fixed frequency clock to the digital conversion unit  150 . Each of the ADCs  151  to  154  performs the digital sampling in synchronism with the clock output from the fixed-frequency oscillator  411 . The phase control circuit  162  detects the phases of the respective signals output from the waveform distortion compensation circuit  161  and outputs the frequency control signal and the phase control signal to the frequency/phase compensation circuit  412 . 
         [0087]    The frequency/phase compensation circuit  412  (frequency/phase compensation unit) is provided to the digital signal processing circuit  160 . The frequency/phase compensation circuit  412  performs the frequency compensation and the phase compensation on the signal output from the ADCs  151  to  154  to compensate the sampling phase. To be more specific, on the basis of the frequency control signal and the phase control signal output from the phase control circuit  162 , the frequency/phase compensation circuit  412  compensates the sampling phase of the signal output from the ADCs  151  to  154 . The frequency/phase compensation circuit  412  outputs the signal in which the sampling phase is compensated to the waveform distortion compensation circuit  161 . 
         [0088]    In this manner, the digital coherent receiver  100  performs the frequency compensation and the phase compensation on the signal converted into the digital signal on the basis of the detected phase. With this configuration, it is possible to suppress the influence on the digital processing from the fluctuation of the sampling phase in the digital conversion unit  150 . For this reason, for example, even when such a configuration that the digital conversion unit  150  performs the sampling in synchronism with the clock oscillated by the fixed-frequency oscillator  411  is adopted, it is possible to suppress the influence on the digital processing from the fluctuation of the sampling phase in the digital conversion unit  150 . 
         [0089]    (Specific Example of Phase Control Circuit) 
         [0090]      FIG. 5  is a block diagram showing a specific example 1 of the phase control circuit shown in  FIGS. 1 to 3 . In  FIG. 5 , with regard to a part of the configuration of the digital coherent receiver  100  shown in  FIG. 1 , the respective signals for I and Q channels and the H and V axes are collectively illustrated. As shown in  FIG. 5 , the phase control circuit  162  is provided with a phase adjuster  511  (PHA: PHase Adjuster), a phase detection unit  512  (PD: Phase Detector), a first DLF  513  (Digital Loop Filter), and a second DLF  514 . 
         [0091]    The phase adjuster  511  (phase compensation unit) compensates the phase of the signal output from the waveform distortion compensation circuit  161  on the basis of the phase control signal output from the first DLF  513 . The phase adjuster  511  outputs the signal in which the phase is compensated to the subsequent stage (the adaptive equalization type demodulation circuit  163 ). The phase detection unit  512  detects the phase of the signal output from the phase adjuster  511 . The phase detection unit  512  outputs the detected phase signal indicating the phase to the first DLF  513 . 
         [0092]    The first DLF  513  conducts a signal processing on the phase signal output from the phase detection unit  512 . The signal processing conducted by the first DLF  513  is, for example, noise removal (low pass filter). The first DLF  513  outputs the signal subjected to the signal processing as the phase control signal to the phase adjuster  511 . Also, the first DLF  513  outputs the signal subjected to the signal processing to the second DLF  514 . 
         [0093]    The second DLF  514  performs a signal processing on the signal output from the first DLF  513 . The signal processing performed by the second DLF  514  is, for example, the transform from the phase component into the frequency component. The second DLF  514  outputs the signal subjected to the signal processing as the frequency control signal to the frequency variable oscillator  140 . On the basis of the frequency control signal output from the second DLF  514 , the frequency variable oscillator  140  changes the frequency of the clock to be output. 
         [0094]    In this manner, the phase detection unit  512  is provided to the subsequent stage of the phase adjuster  511  and detects the phase of the signal compensated by the phase adjuster  511 . With this configuration, as the control becomes a feed back control in which the phase compensation result in the phase adjuster  511  returns from the phase detection unit  512  to the phase adjuster  511 , it is possible to easily perform the compensation processing in the phase adjuster  511 . For this reason, the phase compensation in the phase adjuster  511  can be carried out precisely, and it is possible to improve the communication quality. 
         [0095]    In a case where the configuration of the phase control circuit  162  shown in  FIG. 5  is applied to the digital coherent receiver  100  shown in  FIG. 2 , the second DLF  514  outputs the frequency control signal to the DDS  212 . On the basis of the frequency control signal output from the second DLF  514 , the DDS  212  changes the frequency of the clock to be generated. 
         [0096]      FIG. 6  is a block diagram showing a specific example 2 of the phase control circuit shown in  FIGS. 1 to 3 . In  FIG. 6 , a configuration similar to the configuration shown in  FIG. 5  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 6 , the phase detection unit  512  may output the detected phase signal indicating the phase to the first DLF  513  and the second DLF  514 . In this case, the second DLF  514  conducts the signal processing on the phase signal output from the phase detection unit  512 . 
         [0097]      FIG. 7  is a block diagram showing a specific example 1 of a phase adjuster. The phase adjuster  511  shown in  FIG. 7  is a specific example of a time domain compensation type phase adjuster. As shown in  FIG. 7 , the phase adjuster  511  is provided with a tap position adjustment selector  710 , a delay elements  721  to  72   n , a tap coefficient calculation unit  730 , multiplication units  741  to  74   n , and an adder unit  750 . 
         [0098]    To the tap position adjustment selector  710 , the signal output from the former stage of the phase adjuster  511  and an integer part obtained by dividing the phase control signal input to the phase adjuster  511  by the sampling period are input. The tap position adjustment selector  710  switches connection paths of the delay elements  721  to  72   n  in accordance with the input integer part. 
         [0099]    For example, the tap position adjustment selector  710  switches the connection paths so that the signal output from the former stage of the phase adjuster  511  (the waveform distortion compensation circuit  161 ) is input to the delay element  721 . Also, the tap position adjustment selector  710  switches the connection path so that the output of the delay element  721  is connected to the input of the delay element  722 , the output of the delay element  722  is connected to the input of the delay element  723 , . . . , and the output of the delay element  72 ( n− 1) is connected to the input of the delay element  72   n.    
         [0100]    Each of the delay elements  721  to  72   n  delays and outputs the input signal. To the tap coefficient calculation unit  730 , a decimal part obtained by dividing the phase control signal input to the phase adjuster  511  by the sampling period (phase less than 1 sample) is input. On the basis of the input decimal part, the tap coefficient calculation unit  730  calculates the respective tap coefficients of the multiplication units  741  to  74   n.    
         [0101]    For example, the tap coefficient calculation unit  730  calculates the tap coefficients by sampling a filtering waveform such as a sinc function by the input phase of the decimal part. Alternatively, the tap coefficient calculation unit  730  decides the respective tap coefficients on the basis of a table in which the decimal parts and the respective tap coefficients are associated with each other. The table in which the decimal parts and the respective tap coefficients are associated with each other is previously stored, for example, in a memory of the digital coherent receiver  100 . The tap coefficient calculation unit  730  respectively outputs the calculated respective tap coefficients to the multiplication units  741  to  74   n.    
         [0102]    To the multiplication units  741  to  74   n , the output signals of the delay elements  721  to  72   n  and the tap coefficients output from the tap coefficient calculation unit  730  are respectively input. Each of the multiplication units  741  to  74   n  multiplies the thus input output signal by the tap coefficient to be output to the adder unit  750 . The adder unit  750  adds the respective output signals output from the multiplication units  741  to  74   n  to be output to the subsequent stage. 
         [0103]    In a case where the signals are input in N parallel with respect to the phase adjuster  511 , the delay elements  721  to  72   n  are omitted, and input selectors at a maximum width of the integer part of the phase control signal and FIR (Finite Impulse Response) filters having the same tap coefficient calculated by the tap coefficient calculation unit  730  are operated in parallel by N pieces. In this case, the input selectors are provided by the number of the taps. 
         [0104]      FIG. 8  is a block diagram showing a specific example 2 of the phase adjuster. The phase adjuster  511  shown in  FIG. 8  is a specific example of a frequency domain compensation type phase adjuster. As shown in  FIG. 8 , the phase adjuster  511  is provided with a Fourier transform unit  811 , a rotator transform unit  812 , a multiplication unit  813 , and an inverse Fourier transform unit  814 . The Fourier transform unit  811  subjects the signal input to the phase adjuster  511  to Fourier Transform (FFT: Fast Fourier Transform) to be transformed into the frequency domain. The Fourier transform unit  811  outputs the signal subjected to the Fourier transform to the multiplication unit  813 . 
         [0105]    The rotator transform unit  812  performs the rotator transform processing on the phase control signal output from the first DLF  513  and outputs the phase shift coefficient obtained through the rotator transform processing to the multiplication unit  813 . The multiplication unit  813  multiplies the signal output from the Fourier transform unit  811  by the phase shift coefficient output from the rotator transform unit  812  and outputs the multiplied signal to the inverse Fourier transform unit  814 . The inverse Fourier transform unit  814  subjects the signal output from the multiplication unit  813  to Inverse Fourier Transformation (IFFT: Inverse FFT: Inverse Fast Fourier Transformation) to be output to the subsequent stage (the adaptive equalization type demodulation circuit  163 ). 
         [0106]      FIG. 9  is a block diagram showing a specific example of the first DLF. As shown in  FIG. 9 , the first DLF  513  is provided with a low pass filter  911  (LPF: Low Pass Filter), a multiplication circuit  912 , an adder circuit  913 , a delay element  914 , a multiplication circuit  915 , a low pass filter  916 , and an adder circuit  917 . To the low pass filter  911 , the phase signal output from the phase detection unit  512  is input. The low pass filter  911  extracts a low frequency component of the input phase signal and outputs the extracted signal to the multiplication circuit  912  and the multiplication circuit  915 . 
         [0107]    The multiplication circuit  912  multiplies the signal output from the low pass filter  911  by a coefficient b to be output to the adder circuit  913 . The adder circuit  913  adds the signal output from the multiplication circuit  912  with the signal output from the delay element  914  to output the added signal as an integral term to the delay element  914  and the adder circuit  917 . The delay element  914  delays the signal output from the adder circuit  913  by one operational clock of the first DLF and outputs the delayed signal to the adder circuit  913 . 
         [0108]    The multiplication circuit  915  multiplies the signal output from the low pass filter  911  by a coefficient a to be output to the low pass filter  916 . The low pass filter  916  extracts the low frequency component of the signal output from the multiplication circuit  915  and outputs the extracted signal as a proportional term to the adder circuit  917 . The adder circuit  917  adds the signal of the integral term output from the adder circuit  913  with the signal of the proportional term output from the low pass filter  916 . The adder circuit  917  outputs the added signal as the phase control signal to the phase adjuster  511 . 
         [0109]    With the configuration, the phase signal input to the first DLF  513  is converted as a sum of the proportional term and the integral term having the coefficients a and b into the phase control signal. The coefficients a and b are decided, for example, in accordance with a design and a transmission mode of the digital coherent receiver  100 . 
         [0110]    The low pass filter  911  operates as a decimation filter for processing the respective phase signals of the paralleled respective signals (the I and Q channels and the H and V axes). For example, as a simple example, the low pass filter  911  outputs an average or a total sum of the respective phase signals. it is also possible to adopt a configuration omitting the low pass filter  911 . 
         [0111]    The low pass filter  916  is provided for suppressing the high frequency noise component of the phase signal. In some cases, the frequency fluctuation of the local light source  112  may have a component equal to or higher than several hundreds of kHz. For this reason, in order to minimize the control loop delay, the low pass filter  916  for suppressing the high frequency noise is inserted only to the proportional term. it is also possible to adopt a configuration omitting the low pass filter  916 . 
         [0112]      FIG. 10  is a block diagram showing a specific example of the second DLF. As shown in  FIG. 10 , the second DLF  514  is provided with a multiplication circuit  1011 , an adder circuit  1012 , a delay element  1013 , a multiplication circuit  1014 , an adder circuit  1015 , and a low pass filter  1016 . The phase signal input to the second DLF  514  (or the phase control signal input) is input to the multiplication circuit  1011  and the multiplication circuit  1014 . 
         [0113]    The multiplication circuit  1011  multiplies the input signal by a coefficient B to be output to the adder circuit  1012 . The adder circuit  1012  adds the signal output from the multiplication circuit  101  with the signal output from the delay element  1013  and outputs the added signal as the integral term to the delay element  1013  and the adder circuit  1015 . The delay element  1013  delays the signal output from the adder circuit  1012  by one operational clock of the second DLF and outputs the delayed signal to the adder circuit  1012 . 
         [0114]    The multiplication circuit  1014  multiplies the input signal by a coefficient A and outputs the multiplied signal as the proportional term to the adder circuit  1015 . The adder circuit  1015  adds the signal of the integral term output from the adder circuit  1012  with the signal of the proportional term output from the multiplication circuit  1014  to be output to the low pass filter  1016 . The low pass filter  1016  extracts the low frequency component of the signal output from the adder circuit  1015  and outputs the extracted signal as the frequency control signal to the frequency variable oscillator  140 . 
         [0115]    With the configuration, the signal input to the second DLF  514  is converted as a sum of the proportional term and the integral term having the coefficients A and B into the frequency control signal. The coefficients A and B are decided, for example, in accordance with a design and a transmission mode of the digital coherent receiver  100 . 
         [0116]    For example, as shown in  FIG. 6 , in a case where the phase signal output from the phase detection unit  512  is directly input to the second DLF  514 , a low pass filter may be provided in the former stage of the multiplication circuit  1011  and the multiplication circuit  1014 . The low pass filter provided in the former stage of the multiplication circuit  1011  and the multiplication circuit  1014  conducts the integral operation of the decimation filter for the phase signals and the phase information. Also, the low pass filter  1016  is a low pass filter for avoiding the high frequency noise placed on the clock output from the frequency variable oscillator  140 . it is also possible to adopt a configuration omitting the low pass filter  1016 . 
         [0117]      FIG. 11  is a block diagram showing a specific example 3 of the phase control circuit shown in  FIGS. 1 to 3 . In  FIG. 11 , a configuration similar to the configuration shown in  FIG. 5  is assigned with the same reference symbol, and a description thereof is omitted. In a case where the waveform distortion compensation circuit  161  is a circuit for conducting the waveform distortion compensation in the frequency domain, as shown in  FIG. 11 , a compensation circuit  1111  in which the waveform distortion compensation circuit  161  and the phase adjuster  511  are integrally constructed may be provided instead of the waveform distortion compensation circuit  161  and the phase adjuster  511  shown in  FIG. 5 . such a configuration may be adopted that instead of the frequency variable oscillator  140  shown in  FIG. 11 , the fixed-frequency oscillator  211  and the DDS  212  (see  FIG. 2 ) are provided. 
         [0118]      FIG. 12  is a block diagram showing a specific example 4 of the phase control circuit shown in  FIGS. 1 to 3 . In  FIG. 12 , a configuration similar to the configuration shown in  FIG. 6  is assigned with the same reference symbol, and a description thereof is omitted. In a case where the waveform distortion compensation circuit  161  is a circuit for conducting the waveform distortion compensation in the frequency domain, as shown in  FIG. 12 , instead of the waveform distortion compensation circuit  161  and the phase adjuster  511  shown in  FIG. 6 , the compensation circuit  1111  in which the waveform distortion compensation circuit  161  and the phase adjuster  511  are integrally constructed may be provided. such a configuration may be adopted that instead of the frequency variable oscillator  140  shown in  FIG. 12 , the fixed-frequency oscillator  211  and the DDS  212  (see  FIG. 2 ) are provided. 
         [0119]      FIG. 13  is a block diagram showing a specific example 1 of the compensation circuit. The compensation circuit  1111  shown in  FIGS. 11 and 12  is provided, for example, as shown in  FIG. 13 , with a Fourier transform unit  1311 , a rotator transform unit  1312 , a multiplication unit  1313 , a multiplication unit  1314 , and an inverse Fourier transform unit  1315 . 
         [0120]    The Fourier transform unit  1311  performs the Fourier transform on the signal input to the compensation circuit  1111  to be transformed into the frequency domain. The Fourier transform unit  1311  outputs the signal subjected to the Fourier transform to the multiplication unit  1313 . The rotator transform unit  1312  performs the rotator transform processing on the phase control signal output from of the first DLF  513  and outputs the phase shift coefficient obtained through the rotator transform processing to the multiplication unit  1314 . 
         [0121]    The multiplication unit  1313  multiplies the signal output from the Fourier transform unit  1311  by the waveform distortion correction coefficient in the frequency domain and outputs the multiplied signal to the multiplication unit  1314 . The waveform distortion correction coefficient multiplied in the multiplication unit  1313  is a coefficient decided in accordance with the waveform distortion of the reception signal and is previously stored, for example, in the memory of the digital coherent receiver  100 . 
         [0122]    The multiplication unit  1314  multiplies the signal output from the multiplication unit  1313  by the phase shift coefficient output from the rotator transform unit  1312  and outputs the multiplied signal to the inverse Fourier transform unit  1315 . The inverse Fourier transform unit  1315  subjects the signal output from the multiplication unit  1314  to the inverse Fourier transform to be output to the subsequent stage (the adaptive equalization type demodulation circuit  163 ). such a configuration may also be adopted that the multiplication unit  1314  is provided in the former stage of the multiplication unit  1313 . That is, the order for multiplying the waveform distortion correction coefficient and the phase shift coefficient does not make difference in either way. 
         [0123]    In this manner, the waveform distortion compensation circuit  161  and the phase adjuster  511  can be realized by the compensation circuit  1111  for multiplying the waveform distortion correction coefficient by the phase shift coefficient obtained by transforming the phase control signal transformed by the first DLF  513  into the rotator of the respective frequencies in the frequency domain. With this configuration, the waveform compensation and the phase compensation can be carried out by performing the Fourier transform by one time. For this reason, it is possible to realize the miniaturization and speeding up of the circuit. 
         [0124]      FIG. 14  is a block diagram showing a specific example 2 of the compensation circuit. In  FIG. 14 , a configuration similar to the configuration shown in  FIG. 13  is assigned with the same reference symbol, and a description thereof is omitted. The waveform distortion compensation circuit  161  is a dispersion compensator for compensating the wavelength dispersion of the signal, and in a case where the waveform distortion compensation target in the frequency domain is the wavelength dispersion, the compensation circuit  1111  may have a configuration shown in  FIG. 14 . Herein, the compensation circuit  1111  has a configuration omitting the multiplication unit  1313  shown in  FIG. 13 . 
         [0125]    The Fourier transform unit  1311  outputs the signal subjected to the Fourier transform to the multiplication unit  1314 . The rotator transform unit  1312  (rotator transformer) conducts the rotator transform processing at the wavelength dispersion compensation amount with the phase control signal output from the first DLF  513  and outputs the rotator obtained through the rotator transform processing (the wavelength dispersion and the shift coefficient of the phase) to the multiplication unit  1314 . The wavelength dispersion amount at which the rotator transform processing is conducted in the rotator transform unit  1312  is a coefficient decided in accordance with the wavelength dispersion of the reception signal and is previously stored, for example, in the memory of digital coherent receiver  100 . 
         [0126]    The multiplication unit  1314  multiplies the Fourier transform unit  1311  by the rotator output from the rotator transform unit  1312  and outputs the multiplied signal to the inverse Fourier transform unit  1315 . In this manner, with use of a state in which the wavelength dispersion compensation coefficient represented by the expression (6) has an amplitude of 1.0 and has only phase angle information, by performing the rotator transform of the wavelength dispersion amount with the phase angle information on the phase shift coefficient of the phase compensation processing, the multiplication in the frequency domain can be conducted by one time. 
         [0127]    The processing of the rotator transform unit  1312  can be represented, for example, by the following expression (9). In the following expression (9), Δτ denotes a phase control amount in the time domain. 
         [0000]    
       
         
           
             
               
                 
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         [0128]    In this manner, in a case where the waveform distortion compensation circuit  161  compensates the wavelength dispersion, the compensation circuit  1111  is provided with the rotator transform unit  1312  for transforming the wavelength dispersion compensation amount and the phase control signal into rotators having the respective frequencies. Then, as the compensation circuit  1111  multiplies the signal by the rotator transformed by the rotator transform unit  1312 , it is possible to carry out the waveform compensation and the phase compensation by conducting the complex multiplication by one time. For this reason, it is possible to realize the miniaturization and speeding up of the circuit. 
         [0129]      FIG. 15  is a block diagram showing a specific example 1 of the phase control circuit shown in  FIG. 4 . In  FIG. 15 , a configuration similar to the configuration shown in  FIG. 5  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 15 , the second DLF  514  outputs the signal subjected to the signal processing as the frequency control signal to the frequency/phase compensation circuit  412 . 
         [0130]    On the basis of the frequency control signal output from the second DLF  514 , the frequency/phase compensation circuit  412  compensates the sampling phase of the signal from the digital conversion unit  150 . The frequency/phase compensation circuit  412  outputs the signal in which the sampling phase is compensated to the waveform distortion compensation circuit  161 . The waveform distortion compensation circuit  161  compensates the waveform distortion of the signal from the frequency/phase compensation circuit  412 . 
         [0131]    In the configuration shown in  FIG. 15 , such a configuration may be adopted that instead of the waveform distortion compensation circuit  161  and the phase adjuster  511 , the compensation circuit  1111  in which the waveform distortion compensation circuit  161  and the phase adjuster  511  are integrally constructed (see  FIGS. 11 to 14 ). 
         [0132]      FIG. 16  is a block diagram showing a specific example 2 of the phase control circuit shown in  FIG. 4 . In  FIG. 16 , a configuration similar to the configuration shown in  FIG. 15  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 16 , the phase detection unit  512  may output the detected phase signal indicating the phase to the first DLF  513  and the second DLF  514 . In this case, the second DLF  514  conducts the signal processing on the phase signal output from the phase detection unit  512 . 
         [0133]    In the configuration shown in  FIG. 16 , such a configuration may be adopted that instead of the waveform distortion compensation circuit  161  and the phase adjuster  511 , the compensation circuit  1111  in which the waveform distortion compensation circuit  161  and the phase adjuster  511  are integrally constructed is provided (see  FIGS. 11 to 14 ). 
         [0134]      FIG. 17  is a block diagram showing a specific example 3 of the phase control circuit shown in  FIG. 4 . In  FIG. 17 , a configuration similar to the configuration shown in  FIG. 15  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 17 , the phase control circuit  162  may have a configuration omitting the phase adjuster  511  in the configuration shown in  FIG. 15 . The first DLF  513  outputs the signal subjected to the signal processing as the phase control signal to the frequency/phase compensation circuit  412 . 
         [0135]    The frequency/phase compensation circuit  412  performs the compensation for the sampling phase on the basis of the frequency control signal from the second DLF  514  and also compensates the phase of the signal output from the waveform distortion compensation circuit  161  on the basis of the phase control signal output from the first DLF  513 . The frequency/phase compensation circuit  412  outputs the signal subjected to the compensation to the waveform distortion compensation circuit  161 . In this manner, the compensation is conducted in the former stage of the waveform distortion compensation circuit  161  also with the inclusion of the phase fluctuation generated in the waveform distortion compensation circuit  161 . 
         [0136]      FIG. 18  is a block diagram showing a specific example 4 of the phase control circuit shown in  FIG. 4 . In  FIG. 18 , a configuration similar to the configuration shown in  FIG. 17  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 18 , the phase detection unit  512  may output the detected phase signal indicating the phase to the first DLF  513  and the second DLF  514 . In this case, the second DLF  514  conducts the signal processing on the phase signal output from the phase detection unit  512 . 
         [0137]      FIG. 19  is a block diagram showing a specific example 1 of the frequency/phase compensation circuit. The frequency/phase compensation circuit  412  shown in  FIG. 19  is a specific example of a time domain compensation type digital frequency/phase compensation circuit. Herein, it is assumed that the oscillation frequency of the fixed-frequency oscillator  411  is set slightly higher than the reception signal. As shown in  FIG. 19 , the frequency/phase compensation circuit  412  is provided with a frequency phase converter  1910 , a parallel conversion unit  1920 , a tap coefficient calculation unit  1930 , and N pieces of FIR filters  1940 . 
         [0138]    The frequency phase converter  1910  converts the output of the second DLF  514  from the frequency into the phase for using the output of the second DLF  514  (the frequency control signal) as the phase control signal. The frequency phase converter  1910  is, for example, an integrator. An integer part of the signal converted by the frequency phase converter  1910  into the phase is output to the parallel conversion unit  1920  and also deducted as the number of samples where the control is ended in the frequency phase converter  1910 . 
         [0139]    On the basis of the integer part of the signal output from the frequency phase converter  1910 , the parallel conversion unit  1920  converts the signal input to the frequency/phase compensation circuit  412  into the parallel signal. To be more specific, by using the integer part output from the frequency phase converter  1910  as the control signal, the parallel conversion unit  1920  performs the parallel conversion of 1 to N (in a case where the integer part is “0”) or 1 to N+1 (in a case where the integer part is “1”) to be output to the subsequent stage. 
         [0140]    M pieces (N−1−M) to (N−1) of previous time latest data denoted by reference numeral  1921  is held in a case where the integer part of the output from the frequency phase converter  1910  is “0”. Also, M pieces (N−M) to N of the previous time latest data is held in a case where the integer part of the output from the frequency phase converter  1910  is “1” as the parallel conversion of 1 to N+1 is carried out in the parallel conversion unit  1920 . 
         [0141]    Also, the parallel conversion unit  1920  generates a clock for performing the signal processing in the subsequent stage of the parallel conversion unit  1920 . To be more specific, the parallel conversion unit  1920  generates clocks of 1/N (in a case where the integer part is “0”) or 1/(N+1) (in a case where the integer part is “1”) of the sampling clocks of the digital conversion unit  150  to be output to the subsequent stage. In a case where the parallel conversion of 1 to N+1 is performed, the parallel conversion unit  1920  creates clocks so that one clock time of the subsequent stage of the parallel conversion unit  1920  becomes the N+1 sample time. 
         [0142]    The decimal part converted into the phase by the frequency phase converter  1910  is output to the tap coefficient calculation unit  1930 . On the basis of the decimal part of the output from the frequency phase converter  1910 , the tap coefficient calculation unit  1930  calculates the respective tap coefficients which become the sample positions for the N pieces of FIR filters  1940  (0 to N−1). The tap coefficient calculation unit  1930  outputs the calculated respective tap coefficients to the respectively corresponding FIR filters  1940 . The processing by the tap coefficient calculation unit  1930  includes a latency adjustment equivalent to the parallel conversion unit  1920 . 
         [0143]    For example, in a case where the frequency difference between the reception signal and the fixed-frequency oscillator  411  is small, the tap coefficients with respect to the N pieces of FIR filters  1940  may be set to be identical. 
         [0144]    Each of the N pieces of FIR filters  1940  (0 to N−1) compensates the respective signals output from the parallel conversion unit  1920  on the basis of the tap coefficients output from the tap coefficient calculation unit  1930 . Each of the FIR filters  1940  (0 to N−1) outputs the compensated signal to the subsequent stage as N sample paralleled data. 
         [0145]    Also, as in the configuration shown in  FIGS. 17 and 18 , in a case where the output of the first DLF  513  (the phase control signal) is also input to the frequency/phase compensation circuit  412 , an adder circuit  1950  for adding the output of the first DLF  513  to the output of the frequency phase converter  1910  may be provided. 
         [0146]      FIG. 20  is a block diagram showing a specific example 2 of the frequency/phase compensation circuit. In  FIG. 20 , a configuration similar to the configuration shown in  FIG. 19  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 20 , instead of the tap coefficient calculation unit  1930  and the FIR filters  1940  shown in  FIG. 19 , the frequency/phase compensation circuit  412  may be provided with a Fourier transform unit  2011 , a rotator transform unit  2012 , a multiplication unit  2013 , and an inverse Fourier transform unit  2014 . 
         [0147]    The parallel conversion unit  1920  outputs the parallel data (N+1 data) subjected to the parallel conversion to the Fourier transform unit  2011 . The Fourier transform unit  2011  subjects the signal output from the parallel conversion unit  1920  to the Fourier transform to be transformed into the frequency domain. To be more specific, in a case where the integer part of the output from the frequency phase converter  1910  is “0”, the Fourier transform unit  2011  performs the processing by only using the 1 to N-th inputs. 
         [0148]    Also, in a case where the integer part of the output from the frequency phase converter  1910  is “1” and already the FFT segment begins, the Fourier transform unit  2011  uses the 1 to (N+1)-th inputs to be input to the FFT as the continuous sample. Then, until the FFT segment ends, the Fourier transform unit  2011  uses all the signals output from the parallel conversion unit  1920 . The last FFT inputs are the 1 to (N−1)-th. 
         [0149]    In a case where the FFT segment is to begin after this, the Fourier transform unit  2011  uses the 2 to (N+1)-th inputs to start the FFT and thereafter uses the 1 to N-th inputs. In a case where the FFT segment is about to end, the Fourier transform unit  2011  uses the 1 to N-th inputs, and the FFT window ends. The Fourier transform unit  2011  outputs the signal subjected to the Fourier transform to the multiplication unit  2013 . 
         [0150]    The rotator transform unit  2012  performs the rotator transform processing on the decimal part of the output from the frequency phase converter  1910  and outputs the shift coefficient obtained through the rotator transform processing to the multiplication unit  2013 . The processing by the rotator transform unit  2012  includes a latency adjustment equivalent to the parallel conversion unit  1920  and the Fourier transform unit  2011 . 
         [0151]    The multiplication unit  2013  multiplies the signal output from the Fourier transform unit  2011  by the shift coefficient output from the rotator transform unit  2012  and outputs the multiplied signal to the inverse Fourier transform unit  2014 . The inverse Fourier transform unit  2014  subjects the signal output from the multiplication unit  2013  to the inverse Fourier transform to be output to the subsequent stage (the waveform distortion compensation circuit  161 ). 
         [0152]    A phase shift of a decimal part Δτ of the output from the frequency phase converter  1910  becomes a rotator coefficient exp(jωΔτ) in the frequency domain. For this reason, the Fourier transform result of the input signal is multiplied by the rotator coefficient to conduct the inverse Fourier transform so that the phase shift is realized, the frequency domain processing in the Fourier transform unit  2011 , the multiplication unit  2013 , and the inverse Fourier transform unit  2014  can be commonly used not only as the frequency/phase compensation but also, for example, as the compensation processing for the wavelength dispersion. 
         [0153]      FIG. 21  is a block diagram showing a specific example 5 of the phase control circuit shown in  FIGS. 1 to 3 . In  FIG. 21 , a configuration similar to the configuration shown in  FIG. 5  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 21 , the phase control circuit  162  may be provided with a phase detection unit  2111  in addition to the configuration shown in  FIG. 5 . The phase detection unit  512  detects the phase of the signal output from the waveform distortion compensation circuit  161  to the phase adjuster  511 . The phase detection unit  512  outputs the detected phase signal indicating the phase to the first DLF  513 . 
         [0154]    The phase detection unit  2111  detects the phase of the signal output from the phase adjuster  511 . The phase detection unit  2111  outputs the detected phase signal indicating the phase to the second DLF  514 . The first DLF  513  conducts the signal processing on the phase signal output from the phase detection unit  512  and outputs the signal subjected to the signal processing to the phase adjuster  511 . The second DLF  514  conducts the signal processing on the phase signal output from the phase detection unit  2111 . The second DLF  514  outputs the signal subjected to the signal processing as the frequency control signal to the frequency variable oscillator  140 . 
         [0155]    In this manner, the phase detection unit  512  may have a configuration of detecting the phase of the signal before being compensated by the phase adjuster  511  in the configuration shown in  FIG. 5 . In this case, the control becomes a feed forward control in which the phase detection result by the phase detection unit  512  is output to the subsequent stage of the phase adjuster  511 . such a configuration may be adopted that instead of the frequency variable oscillator  140  shown in  FIG. 21 , the fixed-frequency oscillator  211  and the DDS  212  (see  FIG. 2 ) are provided. 
         [0156]      FIG. 22  is a block diagram showing a specific example 6 of the phase control circuit shown in  FIGS. 1 to 3 . In  FIG. 22 , a configuration similar to the configuration shown in  FIG. 6  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 22 , the phase detection unit  512  of the phase control circuit  162  detects the phase of the signal output from the waveform distortion compensation circuit  161  to the phase adjuster  511 . 
         [0157]    In this manner, the phase detection unit  512  may have a configuration of detecting the phase of the signal before being compensated by the phase adjuster  511  in the configuration shown in  FIG. 6 . In this case, the control becomes the feed forward control in which the phase detection result by the phase detection unit  512  is output to the subsequent stage of the phase adjuster  511 . such a configuration may be adopted that instead of the frequency variable oscillator  140  shown in  FIG. 22 , the fixed-frequency oscillator  211  and the DDS  212  (see  FIG. 2 ) are provided. 
         [0158]    (Configuration Example of Phase Detection Unit) 
         [0159]      FIG. 23  is a block diagram showing a configuration example of a phase detector used for the phase detection unit  512 . A phase detector  2300  shown in  FIG. 23  is a Gardner-system phase detector (for example, see F. M. Gardner, “A BPSK/QPSK timing-error detector for sampled receivers” mentioned above). As shown in  FIG. 23 , the phase detector  2300  is provided with a delay element  2311 , a delay element  2312 , a subtraction unit  2313 , a multiplication unit  2314 , a delay element  2321 , a delay element  2322 , a subtraction unit  2323 , a multiplication unit  2324 , and an adder unit  2330 . To the phase detector  2300 , for example, a signal subjected to 2× over sampling is input. 
         [0160]    An I channel component (H_i or V_i) of the signal input to the phase detector  2300  is input to the delay element  2311  and the subtraction unit  2313 . The delay element  2311  delays the input signal by ½ symbols and outputs the delayed signal to the delay element  2312  and the multiplication unit  2314 . The delay element  2312  delays the signal output from the delay element  2311  by ½ symbols to be output to the subtraction unit  2313 . 
         [0161]    The subtraction unit  2313  subtracts the signal input to the phase detector  2300  from the signal output from the delay element  2312  to be output to the multiplication unit  2314 . The signal output from the subtraction unit  2313  is a difference between the signals deviated by 1 symbol. The multiplication unit  2314  multiplies the ½ symbols deviated signal output from the delay element  2311  by a difference between the 1 symbol deviated signals output from the subtraction unit  2313  to be output to the adder unit  2330 . 
         [0162]    A Q channel component (H_q or V_q) of the signal input to the phase detector  2300  is input to the delay element  2321  and the subtraction unit  2323 . The delay element  2321  delays the input signal by ½ symbols and outputs the delayed signal to the delay element  2322  and the multiplication unit  2324 . The delay element  2322  delays the signal output from the delay element  2321  by ½ symbols to be output to the subtraction unit  2323 . 
         [0163]    The subtraction unit  2323  subtracts the signal input to the phase detector  2300  from the signal output from the delay element  2322  to be output to the multiplication unit  2324 . The signal output from the subtraction unit  2323  is a difference between the 1 symbol deviated signals. The multiplication unit  2324  multiplies the ½ symbols deviated signal output from the delay element  2321  by a difference between the 1 symbol deviated signals output from the subtraction unit  2323  to be output to the adder unit  2330 . 
         [0164]    The adder unit  2330  adds the signal output from the multiplication unit  2314  with the signal output from the multiplication unit  2324  to be output to the subsequent stage. The processing in the adder unit  2330  is conducted on the basis of a symbol rate (=½ down sampling). With this configuration, the signal output from the adder unit  2330  becomes a phase signal where the signal of the ½ symbols deviated phase is a 0 cross point. 
         [0165]    Herein, it is also conceivable that the Gardner system phase detector  2300  shown in  FIG. 23  can be used as the phase detection unit  512 , the phase detection sensitivity changes because of the wavelength dispersion compensation error (SD) and the polarized wave mode dispersion shown in the expressions (6) and (7). In particular, the change in phase detection sensitivity based on the polarized wave mode dispersion has a dependency with respect to the polarized wave rotation state of the optical fiber. 
         [0166]      FIG. 24  is a graph showing a sensitivity correction by a phase detector of a sensitivity correction type (single-sided correction). In  FIG. 24 , the horizontal axis represents a phase of the signal input to the phase detector. The vertical axis represents an amplitude of the phase signal output from the phase detector. A relation 2410 represents a relation between the phase of the signal and the amplitude of the phase signal in a case where the sensitivity degradation in the phase detector does not exist. A relation 2420 represents a relation between the phase of the signal and the amplitude of the phase signal in a case where the sensitivity degradation in the phase detector exists. 
         [0167]    Typically, as shown in the relation 2410, the phase detector linearly detects the phase in a range about ±0.15 to 0.2 symbols by using the 0 cross point as the center. However, the phase detection sensitivity indicated by an inclination of the phase detection result is degraded because of the wavelength dispersion compensation error (ΔD) and the polarized wave mode dispersion represented by the expressions (6) and (7). For this reason, as shown in the relation 2410, the phase detection result has an inclination different from a phase detection result expectation value. 
         [0168]    This sensitivity degradation adversely affects the phase control loop through which the first DLF  513  and the second DLF  514  are inserted. For this reason, a phase shift amount x is set in a range where the phase detector linearly performs the phase detection, and a phase detection result α of the input signal is corrected on the basis of a phase detection result β of the x phase shifted signal (single-sided correction). The correction coefficient is in proportion to 1/(β−α), but as the current phase is close to the origin, the correction coefficient may be in proportion to 1/β. 
         [0169]      FIG. 25  is a graph showing a sensitivity correction by a phase detector of a sensitivity correction type (two-sided correction). In  FIG. 25 , a part similar to the part shown in  FIG. 24  is assigned with the same reference symbol, and a description thereof is omitted. The phase detection result α of the input signal may be corrected on the basis of the phase detection results β and γ of the x and −x phase shifted signals (two-sided correction). The correction coefficient is assumed to be in proportion to 2/(β−γ) as the current phase is close to the origin. A proportionality coefficient for the correction value is decided on the basis of the phase shift amount x and may be decided so as to be the inclination of the phase detection result expectation value through the multiplication of the correction coefficient. Also, β of the single-sided correction and (α−γ) of the two-sided correction may be set as negative values. 
         [0170]      FIG. 26  is a block diagram showing a configuration example of a sensitivity monitor phase detector (single-sided monitor). As shown in  FIG. 26 , a sensitivity monitor phase detector  2600  is provided with a phase detector  2611  and a sensitive monitor unit  2620 . To the sensitive monitor unit  2620 , a branched signal input to the sensitivity monitor phase detector  2600  is input. The phase detector  2611  detects the phase of the input signal and outputs the detected phase signal indicating the phase (α in  FIGS. 24 and 25 ) to the subsequent stage. 
         [0171]    The sensitive monitor unit  2620  is provided with an x phase shift unit  2621  and a phase detector  2622  (a second phase detector). The x phase shift unit  2621  shifts the phase of the input signal by the shift amount x. For example, the x phase shift unit  2621  generates a signal in which the phase is shifted by the shift amount x through an inter-sample interpolation or the like. The x phase shift unit  2621  outputs the phase shifted signal to the phase detector  2622 . 
         [0172]    The phase detector  2622  detects the phase of the signal output from the x phase shift unit  2621 . The phase detector  2622  is a phase detector having a sensitivity degradation characteristic similar to the phase detector  2611 . The phase detector  2622  outputs the detected phase signal indicating the phase as the sensitivity monitor value (β in  FIGS. 24 and 25 ) to the subsequent stage. 
         [0173]    Also, in a case where the parallel signals are input to the sensitivity monitor phase detector  2600 , such a configuration may be adopted that an averaging unit  2612  (Σ) is provided in the subsequent stage of the phase detector  2611 , and the phase signals of the respective signals output from the phase detector  2611  may be averaged by the averaging unit  2612 . Also, in a case where the respective signals on the H axis and the V axis are input to the sensitivity monitor phase detector  2600 , a polarized wave diversity addition may be conducted in the averaging unit  2612 . 
         [0174]    Also, in a case where the parallel signals are input to the sensitivity monitor phase detector  2600 , for example, such a configuration may be adopted that a down sampling unit  2623  is provided in the former stage of the x phase shift unit  2621 , and down sampling is conducted in accordance with a sensitivity fluctuation speed. The sensitive monitor unit  2620  can adopt a configuration of conducting the down sampling as the sensitive monitor unit  2620  may be operated at a speed where the operation can follow a state affecting the phase detection sensitivity (polarized characteristic state fluctuation or the like) among the state fluctuations of the optical transmission path. 
         [0175]    Also, in a case where the parallel signals are input to the sensitivity monitor phase detector  2600 , such a configuration may be adopted that an averaging unit  2624  (Σ) is provided in the subsequent stage of the phase detector  2622 , the phase signals of the respective signals output from the phase detector  2622  may be averaged by the averaging unit  2624 . Also, in a case where the respective signals on the H axis and the V axis are input to the sensitivity monitor phase detector  2600 , the polarized wave diversity addition may be conducted in the averaging unit  2624 . Also, such a configuration may be adopted that a low pass filter  2625  is provided in the output stage of the sensitive monitor unit  2620 , and wide-area noise of the sensitivity monitor value is suppressed. 
         [0176]    In this manner, the sensitive monitor unit  2620  shifts the phase of the signal and detects the phase of the phase shifted signal, so that it is possible to monitor the detection sensitivity of the phase detector  2611 . Also, the x phase shift unit  2621  shifts the phase in a range where the phase detector  2611  linearly detects the phase. With this configuration, the detection sensitivity of the phase detector  2611  can be monitored accurately. 
         [0177]      FIG. 27  is a block diagram showing a configuration example of a sensitivity monitor phase detector (two-sided monitor). In  FIG. 27 , a configuration similar to the configuration shown in  FIG. 26  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 27 , the sensitive monitor unit  2620  of the sensitivity monitor phase detector  2600  is provided with a −x phase shift unit  2711  (a second phase shift unit), a phase detector  2712  (a third phase detector), and a subtraction unit  2713  in addition to the configuration shown in  FIG. 26 . 
         [0178]    The phase detector  2622  outputs the phase signal to the subtraction unit  2713 . The −x phase shift unit  2711  shifts the phase of the input signal by a shift amount −x (opposite direction of the shift amount x). For example, the −x phase shift unit  2711  generates a signal in which the phase is shifted by the shift amount −x through the inter-sample interpolation or the like. The −x phase shift unit  2711  outputs the phase shifted signal to the phase detector  2712 . 
         [0179]    The phase detector  2712  detects the phase of the signal output from the −x phase shift unit  2711 . The phase detector  2712  is a phase detector having a sensitivity degradation characteristic similar to that of the phase detector  2611 . The phase detector  2712  outputs the detected phase signal indicating the phase to the subtraction unit  2713 . The subtraction unit  2713  subtracts the phase signal output from the phase detector  2712  from the phase signal output from the phase detector  2622 . The subtraction unit  2713  outputs the signal indicating the subtraction result as the phase signal to the subsequent stage. 
         [0180]    In this manner, the sensitive monitor unit  2620  calculates a difference between the respective phases of the signal in which the phase is shifted by x and the signal in which the phase is shifted by −x, so that it is possible to monitor the detection sensitivity of the phase detector  2611  with respect to the fluctuations in both directions of the phase. Also, the −x phase shift unit  2711  shifts the phase in a range where the phase detector  2611  linearly detects the phase. With this configuration, it is possible to monitor the detection sensitivity of the sensitivity monitor phase detector  2600  accurately. 
         [0181]      FIG. 28  is a block diagram showing a configuration example of a phase detection unit of a sensitivity selection correction type. As shown in  FIG. 28 , a phase detection unit  2800  is provided with equalization filters  2811  to  281 N, sensitivity monitor phase detectors  2821  to  282 N, a selection unit  2830 , a selection switch  2840 , a sensitivity correction coefficient generation unit  2850 , and a multiplication unit  2860 . The phase detection unit  2800  is a sensitivity selection type phase detection unit and can be applied, for example, to the phase detection unit  512 . 
         [0182]    The equalization filters  2811  to  281 N are equalization filters having mutually different equalization characteristics (multiplication coefficients and the like). To the respective equalization filters  2811  to  281 N, the I channel component (H_i) and the Q channel component (H_q) included in the H axis of the signal and the I channel component (V_i) and the Q channel component (V_q) included in the V axis of the signal are input. The equalization filter  2811  performs an equalization processing on the respective input signals to be output to the sensitivity monitor phase detector  2821 . Similarly, the equalization filters  2812  to  281 N respectively perform the equalization processing on the respective input signals to be respectively output to the sensitivity monitor phase detectors  2822  to  282 N. 
         [0183]    Each of the sensitivity monitor phase detectors  2821  to  282 N is, for example, the sensitivity monitor phase detector  2600  shown in  FIG. 26  or  27 . The sensitivity monitor phase detector  2821  detects the phase of the signal on the basis of the respective signals output from the equalization filter  2811  and outputs the detected phase signal indicating the phase to the selection switch  2840 . Also, the sensitivity monitor phase detector  2821  outputs the sensitive monitor value to the selection unit  2830 . 
         [0184]    Similarly, each of the sensitivity monitor phase detectors  2822  to  282 N detects the phase of the signal on the basis of the respective signals output from the equalization filters  2812  to  281 N and outputs the detected phase signal indicating the phase to the selection switch  2840 . Also, each of the sensitivity monitor phase detectors  2822  to  282 N outputs the sensitive monitor value to the selection unit  2830 . 
         [0185]    On the basis of the sensitivity monitor values output from the sensitivity monitor phase detectors  2821  to  282 N, the selection unit  2830  selects one of the sensitivity monitor phase detectors  2821  to  282 N. To be more specific, the selection unit  2830  selects the sensitivity monitor phase detector that outputs the sensitivity monitor value whose absolute value is largest among the sensitivity monitor phase detectors  2821  to  282 N. In the selection of the sensitivity monitor phase detector, in order to avoid an influence from noise, the detection of the largest value of the sensitivity monitor value may have hysteresis. 
         [0186]    The selection is made on the basis of the absolute value of the sensitivity monitor value because the satisfactory phase detection may be carried out on the basis of a negative sensitivity depending on the polarized wave mode dispersion state of the optical transmission path. The selection unit  2830  notifies the selection switch  2840  of the selected sensitivity monitor phase detector. Also, the selection unit  2830  outputs the largest sensitivity monitor value among the sensitivity monitor values output from the sensitivity monitor phase detectors  2821  to  282 N to the sensitivity correction coefficient generation unit  2850 . 
         [0187]    The selection switch  2840  outputs the phase signal output from the sensitivity monitor phase detector which is notified from the selection unit  2830  among the respective phase signals output from the sensitivity monitor phase detectors  2821  to  282 N to the multiplication unit  2860 . 
         [0188]    The sensitivity correction coefficient generation unit  2850  is provided with an inverse number calculation unit  2851  and a multiplication unit  2852 . The inverse number calculation unit  2851  calculates an inverse number of the sensitivity monitor value output from the selection unit  2830  to be output to the multiplication unit  2852 . The multiplication unit  2852  multiplies the signal output from the inverse number calculation unit  2851  by the coefficient and outputs the multiplication result as the sensitivity correction coefficient to the multiplication unit  2860 . The coefficient multiplied in the multiplication unit  2852  is coefficient equivalent to a phase shift amount x in the sensitivity monitor phase detectors  2821  to  282 N (see  FIG. 26  or  27 ). 
         [0189]    The multiplication unit  2860  multiplies the phase signal output from the selection switch  2840  by the sensitivity correction coefficient output from the multiplication unit  2852 . The multiplication unit  2860  outputs the multiplied phase signal to the subsequent stage. according to the configuration, the sensitivity correction coefficient generation unit  2850  conducts the calculation for the inverse number of the sensitivity monitor value and the coefficient multiplication, but a table reference configuration may be adopted that the sensitivity monitor value is converted into the sensitivity correction coefficient on the basis of a table in which the sensitivity monitor value and the sensitivity correction coefficient are associated with each other. The table in which the sensitivity monitor value and the sensitivity correction coefficient are associated with each other is previously stored, for example, in the memory of digital coherent receiver  100 . 
         [0190]    In this manner, the phase detection unit  2800  performs the equalization processing on the input signals in parallel by the equalization filters  2811  to  281 N having mutually different equalization characteristics and detects the phases of the respective signals subjected to the equalization processing. Also, on the basis of the monitor results of the respective detection sensitivities of the sensitivity monitor phase detectors  2821  to  282 N, the phase detection unit  2800  selects one of the sensitivity monitor phase detectors  2821  to  282 N and outputs the phase signal indicating the phase detected by the selected phase detector. 
         [0191]    With this configuration, the detection result of the phase detector whose detection sensitivity is optimal among the sensitivity monitor phase detectors  2821  to  282 N can be used in the phase adjuster  511 . For example, the detection result of the phase detector whose absolute value of the sensitivity monitor value is largest among the sensitivity monitor phase detectors  2821  to  282 N is used in the phase adjuster  511 . With this configuration, the phase is compensated on the basis of the detection result of the phase detector whose sensitivity degradation is smallest, and it is possible to detect the phase of the signal further accurately. For this reason, it is possible to further improve the communication quality. 
         [0192]    Also, the phase detection unit  2800  generates the sensitivity correction coefficient in proportion to the inverse number of the monitor result of the phase detector selected by the selection unit  2830  among the respective monitor results by the sensitivity monitor phase detectors  2821  to  282 N. Then, the phase detection unit  2800  multiplies the phase output by the selection switch  2840  by the sensitivity correction coefficient. With this configuration, the sensitivity degradation in the selected phase detector is corrected, and it is possible to detect the phase of the signal further accurately. For this reason, it is possible to further improve the communication quality. 
         [0193]      FIG. 29  is a block diagram showing a configuration example 1 of a phase detection unit of a diversity addition type. A phase detection unit  2900  shown in  FIG. 29  is provided with an H axis phase detector  2911  (a first phase detector), a V axis phase detector  2912  (a second phase detector), and an adder unit  2920 . The phase detection unit  2900  is a diversity addition type phase detection unit and can be applied, for example, to the phase detection unit  512 . 
         [0194]    To the H axis phase detector  2911 , the I channel component (H_i) and the Q channel component (H_q) included in the H axis of the signal are input. The H axis phase detector  2911  detects the phase of the input signal and outputs the detected phase signal indicating the phase to the adder unit  2920 . To the V axis phase detector  2912 , the I channel component (V_i) and the Q channel component (V_q) included in the V axis of the signal are input. The V axis phase detector  2912  detects the phase of the input signal and outputs the detected phase signal indicating the phase to the adder unit  2920 . 
         [0195]    The adder unit  2920  adds the phase signal output from the H axis phase detector  2911  with the phase signal output from the V axis phase detector  2912 . The adder unit  2920  outputs the addition result as the phase signal to the subsequent stage. 
         [0196]    In this manner, the phase detection unit  2900  detects the phases of the respective signals on the H axis (the first polarized wave) and the V axis (the second polarized wave) and adds the detected respective phases, so that it is cancel the polarized wave dependency of the phase detection result. Also, it is possible to suppress the noise of the phase detection result. 
         [0197]      FIG. 30  is a block diagram showing a configuration example 2 of the phase detection unit of the diversity addition type. In  FIG. 30 , a configuration similar to the configuration shown in  FIG. 28  is assigned with the same reference symbol, and a description thereof is omitted. A phase detection unit  3000  shown in  FIG. 30  is provided with the equalization filters  2811  to  281 N, phase detectors  3011  to  301 N and  3021  to  302 N, adder units  3031  to  303 N, and a combining unit  3040 . The phase detection unit  3000  is a diversity addition type phase detection unit and can be applied, for example, to the phase detection unit  512 . 
         [0198]    To each of the equalization filters  2811  to  281 N, the I channel component (H_i) and the Q channel component (H_q) included in the H axis of the signal and the I channel component (V_i) and the Q channel component (V_q) included in the V axis of the signal are input. The respective equalization filters  2811  to  281 N perform the equalization processing on the input respective signals. 
         [0199]    The equalization filter  2811  outputs the signal on the H axis subjected to the equalization processing to the phase detector  3011  and the signal on the V axis subjected to the equalization processing to the phase detector  3021 . Similarly, the equalization filters  2812  to  281 N output the signals on the H axis subjected to the equalization processing to the phase detectors  3012  to  301 N, respectively, and the signals on the V axis subjected to the equalization processing to the phase detectors  3022  to  302 N, respectively. 
         [0200]    The phase detector  3011  detects the phase of the signal on the H axis from the equalization filter  2811  and outputs the detected phase signal indicating the phase to the adder unit  3031 . Similarly, the phase detectors  3012  to  301 N respectively detect the phases of the signals on the H axis from the equalization filters  2812  to  281 N and respectively output the detected phase signals indicating the phases to the adder units  3032  to  303 N. 
         [0201]    The phase detector  3021  detects the phase of the signal on the V axis from the equalization filter  2811  and outputs the detected phase signal indicating the phase to the adder unit  3031 . Similarly, the phase detectors  3022  to  302 N respectively detect the phases of the signals on the V axis from the equalization filters  2812  to  281 N and respectively output the detected phase signals indicating the phases to the adder units  3032  to  303 N. 
         [0202]    The adder unit  3031  adds the respective phase signals output from the phase detector  3011  and the phase detector  3021  and outputs the addition result to the combining unit  3040 . Similarly, the adder units  3032  to  303 N respectively adds the respective phase signals output from the phase detectors  3012  to  301 N and the phase detectors  3022  to  302 N and outputs the addition result to the combining unit  3040 . The combining unit  3040  performs the diversity combining of the respective phase signals output from the adder units  3031  to  303 N. The combining unit  3040  outputs the phase signal subjected to the diversity combining to the subsequent stage. 
         [0203]    In this manner, the phase detection unit  3000  performs the diversity addition of the phases detected by the phase detectors  3012  to  301 N and  3022  to  302 N and outputs the addition result as the phase signal. With this configuration, even when the phase detector has the sensitivity degradation, it is possible to detect the phase of the signal accurately. For this reason, as the phase of the signal is compensated accurately, and the digital demodulation in the adaptive equalization type demodulation circuit  163  is conducted accurately, so that it is possible to further improve the communication quality. 
         [0204]      FIG. 31  is a block diagram showing a configuration example 3 of the phase detection unit of the diversity addition type. In  FIG. 31 , a configuration similar to the configuration shown in  FIG. 28  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 31 , a phase detection unit  3100  is provided with a threshold determination unit  3110 , AND circuits  3121  to  312 N, and a combining unit  3130  instead of the selection unit  2830 , the selection switch  2840 , the sensitivity correction coefficient generation unit  2850 , and the multiplication unit  2860  shown in  FIG. 28 . 
         [0205]    The phase detection unit  3100  is a configuration example of the diversity addition type phase detection unit and can be applied, for example, to the phase detection unit  512 . The sensitivity monitor phase detector  2821  outputs the detected phase signal indicating the phase to the AND circuit  3121  and also outputs the sensitive monitor value to the threshold determination unit  3110 . Similarly, the respective sensitivity monitor phase detectors  2822  to  282 N output the detected phase signals indicating the phases to the AND circuits  3122  to  312 N, respectively, and output the sensitive monitor values to the threshold determination unit  3110 . 
         [0206]    The threshold determination unit  3110  conducts a threshold determination on the respective sensitive monitor values output from the sensitivity monitor phase detectors  2821  to  282 N. To be more specific, the threshold determination unit  3110  determines whether or not the sensitive monitor value output from the sensitivity monitor phase detector  2821  exceeds a predetermined threshold and outputs the determination result to the AND circuit  3121 . 
         [0207]    For example, in a case where the sensitive monitor value output from the sensitivity monitor phase detector  2821  exceeds the predetermined threshold, the threshold determination unit  3110  outputs “1” to the AND circuit  3121 , and in a case where the sensitive monitor value is equal to or smaller than the predetermined threshold, the threshold determination unit  3110  outputs “0” to the AND circuit  3121 . Similarly, the threshold determination unit  3110  determines whether or not the sensitivity monitor values from the sensitivity monitor phase detectors  2822  to  282 N exceed a predetermined threshold and outputs the determination results to the respective AND circuits  3122  to  312 N. 
         [0208]    In a case where the determination result output from the threshold determination unit  3110  is “1”, the AND circuit  3121  outputs the phase signal output from the sensitivity monitor phase detector  2821  to the combining unit  3130 . On the other hand, in a case where the determination result output from the threshold determination unit  3110  is “0”, the AND circuit  3121  does not output the phase signal output from the sensitivity monitor phase detector  2821 . 
         [0209]    Similarly, in a case where the determination result output from the threshold determination unit  3110  is “1”, the respective AND circuits  3122  to  312 N output the phase signals respectively output from the sensitivity monitor phase detectors  2822  to  282 N to the combining unit  3130 . On the other hand, in a case where the determination result output from the threshold determination unit  3110  is “0”, the AND circuits  3122  to  312 N do not output the phase signals output from the sensitivity monitor phase detectors  2822  to  282 N. 
         [0210]    The combining unit  3130  performs the diversity combining of the respective phase signals output from the AND circuits  3121  to  312 N. The combining unit  3130  outputs the phase signal subjected to the diversity combining to the subsequent stage. the threshold in the threshold determination unit  3110  may be set as X % of the largest sensitivity monitor value of the sensitivity monitor phase detectors  2821  to  282 N, Y % of the average of the monitor values, or a fixed threshold. 
         [0211]    In this manner, the phase detection unit  3100  monitors the respective detection sensitivities of the sensitivity monitor phase detectors  2821  to  282 N and performs the diversity combining on the phases detected by the phase detectors where it is determined that the monitored respective detection sensitivities exceed the threshold. Then, the phase detection unit  3100  outputs the result of the diversity combining as the phase signal to the subsequent stage. With this configuration, it is possible to exclude the detection result from the phase detector whose sensitivity is significantly degraded, and it is therefore possible to detect the phase of the signal further accurately. For this reason, it is possible to further improve the communication quality. 
         [0212]      FIG. 32  is a block diagram showing a configuration example 4 of the phase detection unit of the diversity addition type. In  FIG. 32 , a configuration similar to the configuration shown in  FIG. 31  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 32 , a phase detection unit  3200  is provided with sensitivity correction coefficient generation units  3211  to  321 N and multiplication units  3221  to  322 N in addition to the configuration shown in  FIG. 31 . 
         [0213]    The phase detection unit  3200  is a configuration example of the diversity addition type phase detection unit and can be applied, for example, to the phase detection unit  512 . The sensitivity monitor phase detector  2821  outputs the sensitivity monitor value to the threshold determination unit  3110  and the sensitivity correction coefficient generation unit  3211 . Similarly, each of the sensitivity monitor phase detectors  2822  to  282 N outputs the sensitivity monitor value to the threshold determination unit  3110  and the sensitivity correction coefficient generation units  3212  to  321 N. 
         [0214]    In a case where the determination result output from the threshold determination unit  3110  is “1”, the AND circuit  3121  outputs the phase signal output from the sensitivity monitor phase detector  2821  to the multiplication unit  3221 . Similarly, in a case where the determination result output from the threshold determination unit  3110  is “1”, the respective AND circuits  3122  to  312 N outputs the phase signals respectively output from the sensitivity monitor phase detectors  2822  to  282 N to the multiplication units  3222  to  322 N, respectively. 
         [0215]    The sensitivity correction coefficient generation unit  3211  generates the sensitivity correction coefficient on the basis of the sensitivity monitor value output from the sensitivity monitor phase detector  2821  and outputs the generated sensitivity correction coefficient to the multiplication unit  3221 . Similarly, on the basis of the sensitivity monitor values respectively output from the sensitivity monitor phase detectors  2822  to  282 N, the sensitivity correction coefficient generation units  3212  to  321 N generate the sensitivity correction coefficients and respectively output the sensitivity correction coefficients to the multiplication units  3222  to  322 N. Each of the sensitivity correction coefficient generation units  3211  to  321 N has a configuration, for example, similar to the sensitivity correction coefficient generation unit  2850  shown in  FIG. 28 . 
         [0216]    The multiplication unit  3221  multiplies the phase signal output from the AND circuit  3121  by the sensitivity correction coefficient output from the sensitivity correction coefficient generation unit  3211 . The multiplication unit  3221  outputs the multiplied phase signal to the combining unit  3130 . Similarly, the multiplication units  3222  to  322 N multiply the phase signals respectively output from the AND circuits  3122  to  312 N by the sensitivity correction coefficients respectively output from the sensitivity correction coefficient generation units  3212  to  321 N. The multiplication units  3222  to  322 N output the multiplied phase signals to the combining unit  3130 . The combining unit  3130  performs diversity combining of the respective phase signals output from the multiplication units  3221  to  322 N. The combining unit  3130  outputs the phase signal subjected to the diversity combining to the subsequent stage. 
         [0217]    Also, a configuration in which a divider unit  3240  is provided may be adopted. The threshold determination unit  3110  notifies the divider unit  3240  of a number M of the sensitive monitors where the sensitivity monitor values output from the sensitivity monitor phase detectors  2821  to  282 N exceed the threshold. The combining unit  3130  outputs the phase signal to the divider unit  3240 . The divider unit  3240  divides the phase signal output from the combining unit  3130  by the number M notified from the threshold determination unit  3110  and outputs the division result as the phase signal to the subsequent stage. With this configuration, the detection sensitivity of the phase detection unit  3200  can be set constant. 
         [0218]    In this manner, the phase detection unit  3200  generates a sensitivity correction coefficient in proportion to the inverse number of the monitor result from the phase detector where it is determined that the detection sensitivities exceed the threshold among the sensitivity monitor phase detectors  2821  to  282 N. Then, the phase detection unit  3200  multiplies the respective phases subjected to the diversity addition by the sensitivity correction coefficient. With this configuration, the sensitivity degradation in the phase detector where it is determined that the detection sensitivities exceed the threshold is corrected, and it is possible to detect the phase of the signal further accurately. For this reason, it is possible to further improve the communication quality. 
         [0219]      FIG. 33  is a block diagram showing a specific example of an equalization filter (polarized wave dispersion equalization). For the equalization filters  2811  to  281 N shown in  FIGS. 28 ,  30 ,  31 , and  32 , for example, polarized wave dispersion equalization type equalization filters  2811 ,  2812 , . . . shown in  FIG. 33  can be applied. As shown in  FIG. 33 , the equalization filter  2811  is provided with a polarized wave rotator  3311 , a DGD adder  3321 , and a phase shifter  3331 . 
         [0220]    The polarized wave rotator  3311  rotates the polarized wave axis of the respective signals on the H axis and the V axis input to the equalization filter  2811  and outputs the respective signals where the polarized wave axis is rotated to the DGD adder  3321 . The DGD adder  3321  adds DGD (Differential Group Delay) to the respective signals on the H axis and the V axis output from the polarized wave rotator  3311 . The DGD adder  3321  outputs the respective signals to which the DGD is added to the phase shifter  3331 . 
         [0221]    The phase shifter  3331  shifts the phases of the respective signals on the H axis and the V axis output from the DGD adder  3321  to correct the phase convergent point which may be deviated because of the DGD addition. The phase shifter  3331  outputs the respective signals in which the phases are shifted to the subsequent stage. It is also possible to adopt a configuration omitting the phase shifter  3331 . 
         [0222]    Similarly, the equalization filters  2812  to  281 N are respectively provided with polarized wave rotators  3312  to  331 N, DGD adders  3322  to  332 N, and phase shifters  3332  to  333 N. The polarized wave rotators  3312  to  331 N, the DGD adders  3322  to  332 N, and the phase shifters  3332  to  333 N are respectively similar to the polarized wave rotator  3311 , the DGD adder  3321 , and the phase shifter  3331 , and a description thereof is omitted. 
         [0223]    The polarized wave rotators  3312  to  331 N have mutually different polarized wave rotation amounts. Also, the DGD adders  3321  to  332 N have mutually different DGDs. Also, the phase shifters  3331  to  333 N have mutually different phase shift amounts. With this configuration, the equalization filters  2811  to  281 N have mutually different equalization characteristics. 
         [0224]      FIG. 34  is a block diagram showing a specific example of an equalization filter (wavelength dispersion equalization). For the equalization filters  2811  to  281 N shown in  FIGS. 28 ,  30 ,  31 , and  32 , for example, a filter of a wavelength dispersion equalization type shown in  FIG. 34  can be applied. The equalization filter  2811  is provided with an H axis wavelength dispersion equalizer  3411  and a V axis wavelength dispersion equalizer  3421 . 
         [0225]    The H axis wavelength dispersion equalizer  3411  equalizes the wavelength dispersion of the signal on the H axis input to the equalization filter  2811  and outputs the signal in which the wavelength dispersion is equalized to the subsequent stage. The V axis wavelength dispersion equalizer  3421  equalizes the wavelength dispersion of the signal on the V axis input to the equalization filter  2811  and outputs the signal in which the wavelength dispersion is equalized to the subsequent stage. 
         [0226]    Similarly, the equalization filters  2812  to  281 N are respectively provided with the H axis wavelength dispersion equalizers  3412  to  341 N and V axis wavelength dispersion equalizers  3422  to  342 N. The H axis wavelength dispersion equalizers  3412  to  341 N and the V axis wavelength dispersion equalizers  3422  to  342 N are respectively similar to the H axis wavelength dispersion equalizer  3411  and the V axis wavelength dispersion equalizer  3421 , and a description thereof is omitted. In this manner, the equalization filters  2811  to  281 N have the wavelength dispersion equalizers corresponding to the respective signals on the H axis and the V axis. For the equalization filter, not only the polarized wave dispersion equalization and the wavelength dispersion equalization are independently applied, but also a combination of those can be applied. 
         [0227]    (Modified Example of Digital Coherent Receiver) 
         [0228]      FIG. 35  is a block diagram showing a modified example 1 of the digital coherent receiver. In  FIG. 35 , with regard to a part of a configuration for the modified example 1 of the digital coherent receiver  100  shown in  FIG. 1 , the I and Q channels and the H and V axes are collectively illustrated. In  FIG. 35 , a configuration similar to the configuration shown in  FIG. 5  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 35 , the digital coherent receiver  100  is provided with a frequency compensator  3511  (frequency compensation unit) and a frequency difference detector  3512  (frequency difference detection unit) instead of the phase adjuster  511  shown in  FIG. 5 . 
         [0229]    The digital conversion unit  150  outputs the digitally converted signal to the frequency compensator  3511 . On the basis of the rotation control signal output from the first DLF  513 , the frequency compensator  3511  compensates the frequency of the signal output from the digital conversion unit  150 . The frequency compensator  3511  outputs the signal in which the frequency is compensated to the waveform distortion compensation circuit  161 . The waveform distortion compensation circuit  161  compensates the waveform distortion of the signal output from the frequency compensator  3511 . 
         [0230]    The phase detection unit  512  detects the phase of the signal output from the waveform distortion compensation circuit  161 . The phase detection unit  512  outputs the detected phase signal indicating the phase to the second DLF  514 . The second DLF  514  performs the signal processing on the signal output from the phase detection unit  512  and outputs the signal subjected to the signal processing as the frequency control signal to the frequency variable oscillator  140 . 
         [0231]    The frequency difference detector  3512  detects the frequency difference of the signal output from the waveform distortion compensation circuit  161 . The frequency difference detector  3512  outputs the frequency difference signal indicating the frequency difference between the detected reception light and the local light to the first DLF  513 . The first DLF  513  performs the signal processing on the frequency difference signal output from the frequency difference detector  3512 . The first DLF  513  outputs the signal subjected to the signal processing as the rotation control signal to the frequency compensator  3511 . Such a configuration may be adopted that instead of the frequency variable oscillator  140  shown in  FIG. 35 , the fixed-frequency oscillator  211  and the DDS  212  (see  FIG. 2 ) are provided. 
         [0232]    In this manner, the digital coherent receiver  100  detects the frequency difference between the reception light and the local light received in the subsequent stage of the waveform distortion compensation circuit  161  and compensates the detected frequency difference fluctuation through the frequency compensation in the former stage of the waveform distortion compensation circuit  161  to suppress the phase fluctuation generated in the output of the waveform distortion compensation circuit  161  due to the frequency fluctuation of the local light source  112 , so that it is possible to accurately carry out the digital demodulation in the adaptive equalization type demodulation circuit  163 . For this reason, it is possible to improve the communication quality. 
         [0233]      FIG. 36  is a block diagram showing a modified example 2 of the digital coherent receiver. In  FIG. 36 , a configuration similar to the configuration shown in  FIG. 35  is assigned with the same reference symbol, and a description thereof is omitted. As shown in  FIG. 36 , the frequency difference detector  3512  may detect a frequency difference of the signal in the subsequent stage of the frequency compensator  3511 . Such a configuration may be adopted that instead of the frequency variable oscillator  140  shown in  FIG. 36 , the fixed-frequency oscillator  211  and the DDS  212  (see  FIG. 2 ). 
         [0234]      FIG. 37  is a block diagram showing a specific example of the frequency difference detector. The frequency difference detector  3512  shown in  FIGS. 35 and 36  is provided, for example, as shown in  FIG. 37 , with computation units  3711  to  3713  and  3721  to  3723 , and an adder unit  3730 . With regard to the signal on the H axis input to the frequency difference detector  3512  (set as X), the computation unit  3711  computes X 4 /|X| 4  and computation result to the computation unit  3712 . 
         [0235]    The computation unit  3712  computes arg( ) with respect to the computation result output from the computation unit  3711  to be converted into the phase information and outputs the computation result to the computation unit  3713 . The computation unit  3713  performs a computation of the following expression (10) on the computation result output from the computation unit  3712  and outputs the computation result to the adder unit  3730 . 
         [0000]    
       
         
           
             
               
                 
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                       1 
                       - 
                       
                         Z 
                         
                           
                             - 
                             2 
                           
                            
                           n 
                         
                       
                     
                     
                       4 
                        
                       n 
                     
                   
                   · 
                   
                     1 
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         [0236]    With regard to the signal on the V axis input to the frequency difference detector  3512  (set as X), the computation unit  3721  computes X 4 /|X| 4  and outputs the computation result to the computation unit  3722 . The computation unit  3722  computes arg( ) with respect to the computation result output from the computation unit  3721  to be converted into the phase information and outputs the computation result to the computation unit  3723 . 
         [0237]    The computation unit  3723  performs the computation of the expression (10) on the computation result output from the computation unit  3722  and outputs the computation result to the adder unit  3730 . The adder unit  3730  adds the respective computation results output from the computation unit  3713  and the computation unit  3723  with each other and outputs the addition result as the frequency difference signal to the subsequent stage. 
         [0238]    In the respective signals on the H axis and the V axis which are the inputs of the frequency difference detectors, signals polarized and multiplexed on the transmitter side are mixed without being separated. In this case, when the modulation system is QPSK (Quadrature Phase Shift Keying), the quadrupling is conducted in the computation units  3721  and  3722 , and the modulation signal nπ/4 (n=1, 3, 5, and 7) on the transmission side becomes nπ (n=1, 3, 5, and 7). 
         [0239]    For this reason, even when any rotation is applied in the optical transmission path, among the adjacent samples, as the complex number, the same phase is realized. For this reason, the conversion into the phase information is carried out in the computation units  3712  and  3722 , and the computation is carried out in the computation units  3713  and  3723 , so that it is possible to calculate the phase rotation amount for one sample from the phase rotation amount (1−Z −2n ) among 2n samples subjected to the 2× over sampling. 
         [0240]    Then, through the addition of the H axis and the V axis in the adder unit  3730 , it is possible to detect the frequency difference as the (2×) phase rotation amount. Even when the maximum frequency difference decided in the system is input, n is decided so that the phase rotation amount among the 2n samples falls within −π to π. Also, in the computation of (1−Z −2n ) in the computation units  3713  and  3723 , as the case may be addition of ±2π is conducted so that the result falls within −π to π. 
         [0241]      FIG. 38  is a block diagram showing a specific example of the frequency compensator. The frequency compensator  3511  shown in  FIGS. 35 and 36  is provided, for example, as shown in  FIG. 38 , with an adder unit  3811 , a remainder operation unit  3812 , a delay element  3813 , a computation unit  3814 , a multiplication unit  3815 , and a multiplication unit  3816 . The adder unit  3811  adds the rotation control signal from the first DLF  513  with a signal θ from the delay element  3813  and outputs the addition result to the remainder operation unit  3812 . 
         [0242]    The remainder operation unit  3812  conducts a remainder operation on the signal output from the adder unit  3811  while 2π is set as a divisor. The remainder operation unit  3812  outputs the signal θ of the computation result to the delay element  3813  and the computation unit  3814 . The delay element  3813  delays the signal θ output from the remainder operation unit  3812  by ½ symbols to be output to the adder unit  3811 . 
         [0243]    On the basis of the signal θ output from the remainder operation unit  3812 , the computation unit  3814  computes a rotator e j θ for each sample. The computation unit  3814  outputs the rotator e j θ obtained through the computation to the multiplication unit  3815  and the multiplication unit  3816 . 
         [0244]    The multiplication unit  3815  multiplies the signal on the H axis input to the frequency compensator  3511  (complex number) by the rotator e j θ output from the computation unit  3814 . The multiplication unit  3815  outputs the multiplied signal on the H axis to the subsequent stage. The multiplication unit  3816  multiplies the signal on the V axis input to the frequency compensator  3511  (complex number) by the rotator e j θ output from the computation unit  3814 . The multiplication unit  3816  outputs the multiplied signal on the V axis to the subsequent stage. 
         [0245]    In a case where the signals are input to the frequency compensator  3511  in parallel, in order to process the N samples at the same time, Z −1  in the delay element  3813  is set as Z −N , and a rotator e jm θ is computed in the m-th signal in the computation unit  3814 . Z −N  is equivalent to one clock delay in the signal processing block. 
         [0246]    (Configuration Example of Optical Transmission System) 
         [0247]      FIG. 39  is a block diagram showing a specific example of the optical transmission system. As shown in  FIG. 39 , an optical transmission system  3900  includes a transmitter  3910  and the digital coherent receiver  100 . The transmitter  3910  transmits the optical signal via the optical transmission path including optical fibers  3911  to  3913  and optical amplifiers  3921  and  3922  to the digital coherent receiver  100 . 
         [0248]    In the optical transmission system  3900 , the waveform distortion of the optical signal such as the wavelength dispersion generated in the optical transmission path can be compensated in the digital coherent receiver  100 . For this reason, for the optical transmission path in the optical transmission system  3900 , a configuration without providing a Dispersion Compensating Fiber (DCF) or the like for compensating the wavelength dispersion amount can also be adopted. 
         [0249]    For this reason, cost reduction, space saving, and the like for the apparatus can be realized, and also the optical attenuation amount of the optical signal can be reduced through no provision of the DCF, so that it is possible to reduce the number of the optical amplifiers can be reduced. For this reason, the power consumption. Also, as compared with the optical waveform compensation circuit, the digital waveform compensation circuit and the digital demodulation circuit in the digital coherent receiver  100  is superior in tracking property with respect to the fluctuation of the transmission path distortion. For this reason, the configuration is also useful to the polarized wave multiplex system where a high tracking property is demanded with respect to the polarized wave. 
         [0250]    (Overlap Type Fourier Transform Unit and Inverse Fourier Transform Unit) 
         [0251]    In the Fourier transform units  811 ,  1311 , and  2011  and the inverse Fourier transform units  815 ,  1315 , and  2014  which are shown in  FIGS. 8 ,  13 ,  14 , and  20 , the phase shift of Δτ in the time domain becomes the rotator coefficient of exp (j ωΔτ )  in the frequency domain. For this reason, the Fourier transform result of the input is multiplied by the rotator coefficient, and the inverse Fourier transform is carried out to realize the phase shift. 
         [0252]    However, if an attempt is made to realize the Fourier transform and the inverse Fourier transform by using the normal FFT, IFFT, DFT (Discrete Fourier Transform), or IDFT (Inverse DFT), discontinuous points may be generated as the sample in which the phase is shifted circulates the Fourier transform window after the inverse Fourier transform. Overlap type Fourier transform unit and inverse Fourier transform unit for solving this phenomenon will be described with reference to  FIGS. 40 and 41 . 
         [0253]      FIG. 40  is a block diagram showing specific example of the Fourier transform unit and the inverse Fourier transform unit. For the Fourier transform units  811 ,  1311 , and  2011  and the inverse Fourier transform units  815 ,  1315 , and  2014  which are shown in  FIGS. 8 ,  13 ,  14 , and  20 , for example, a circuit  4000  shown in  FIG. 40  can be applied. The circuit  4000  is provided with an input unit  4011 , a FFT input frame generation unit  4012 , a FFT processing unit  4013 , a characteristic multiplication unit  4014 , an IFFT processing unit  4015 , an IFFT output frame extraction unit  4016 , and an output unit  4017 . 
         [0254]    Herein, the input data is set as 256 parallel signals, and a window side of FFT and IFFT is set as 1024. The input data (time domain: 256 samples) are input to the input unit  4011 . The input unit  4011  buffers the input data thus input and generates a frame composed of 512 samples for once in 2 clocks. 
         [0255]    The input unit  4011  outputs the generated frame to the FFT input frame generation unit  4012 . Also, the input unit  4011  outputs a control signal including a frame generation timing to an internal counter of the respective blocks of the circuit  4000 . In the subsequent stage of the input unit  4011 , the processing is carried out this frame and the frame generation timing in the input unit  4011  are set as the unit. 
         [0256]    The FFT input frame generation unit  4012  joins the one previous 512 sample frames and the current 512 sample frames in the sample frames output from the input unit  4011  to generate a frame composed of 1024 samples. The FFT input frame generation unit  4012  outputs the generated frame to the FFT processing unit  4013 . 
         [0257]    The FFT processing unit  4013  transforms the frame output from the FFT input frame generation unit  4012  into data in the frequency domain. The FFT processing unit  4013  outputs the transformed frame to the characteristic multiplication unit  4014 . The characteristic multiplication unit  4014  respectively multiplies characteristic parameters for each frequency component with respect to the frequency corresponding to the frame output from the FFT processing unit  4013  (for 1024 frequencies). The characteristic parameters are input, for example, from an external area. The characteristic multiplication unit  4014  outputs the multiplied frame to the IFFT processing unit  4015 . 
         [0258]    The IFFT processing unit  4015  transforms the frame output from the characteristic multiplication unit  4014  into data in the time domain. The IFFT processing unit  4015  outputs the transformed frame to the IFFT output frame extraction unit  4016 . In the vicinity of the frame output from the IFFT processing unit  4015 , the discontinuous points are included. 
         [0259]    With regards the frame output from the IFFT processing unit  4015 , the IFFT output frame extraction unit  4016  discards 256 samples in the back and forth, that is, the quarter of the window size each. If the discontinuous points fall within an area for the discarding in the IFFT output frame extraction unit  4016 , the discontinuous points are not generated in the output obtained by joining the 512 samples which are not discarded. The IFFT output frame extraction unit  4016  outputs the processed frame to the output unit  4017 . 
         [0260]    The output unit  4017  cuts the frame output from the IFFT output frame extraction unit  4016  (512 samples output every 2 samples) into 256 samples each per 1 clock to be output as the parallel signal to the subsequent stage. 
         [0261]      FIG. 41  shows an operation of the circuit shown in  FIG. 40 . Reference numeral  4110  in  FIG. 41  denotes input data input to the input unit  4011 . Reference symbol  4120  denotes N-th frame (FFT input frame) input to the FFT processing unit  4013 . Reference symbol  4130  denotes (N+1)-th frame input to the FFT processing unit  4013 . Reference symbol  4140  denotes (N+2)-th frame input to the FFT processing unit  4013 . 
         [0262]    Reference symbol  4150  denotes N-th frame output from the IFFT processing unit  4015  (IFFT output frame). Reference symbol  4160  denotes (N+1)-th frame output from the IFFT processing unit  4015 . Reference symbol  4170  denotes (N+2)-th frame output from the IFFT processing unit  4015 . Reference symbol  4171  is a frame discarded by the IFFT output frame extraction unit  4016 . 
         [0263]    Reference symbol  4180  denotes a frame in which the respective frames denoted by Reference symbols  4150 ,  4160 , and  4170  (except the frame denoted by reference numeral  4171 ) are joined by the IFFT output frame extraction unit  4016 . In this manner, according to the circuit  4000  for conducting the overlap type FFT and IFFT, the generation of the discontinuous points because of the phase shift can be avoided. 
         [0264]    As described above, according to the digital coherent receiver, it is possible to improve the communication quality. 
         [0265]    The reception method described in the present embodiment can be realized while a previously prepared program is executed by a computer such as a personal computer or a work station. This program is recorded on a computer-readable recording medium such as a hard disk, a flexible disk, a CD-ROM, an MO, or a DVD and, and the program is read out from the recording medium by the computer for the execution. Also, this program may be a transmission medium which can be distributed via a network such as the Internet. 
         [0266]    All examples and conditional language recited herein are intended for pedagogical purposes to aid the reader in understanding the invention and the concepts contributed by the inventor to furthering the art, and are to be construed as being without limitation to such specifically recited examples and conditions, nor does the organization of such examples in the specification relate to a showing of the superiority and inferiority of the invention. Although the embodiment(s) of the present invention has(have) been described in detail, it may be understood that the various changes, substitutions, and alterations could be made hereto without departing from the spirit and scope of the invention.