Abstract:
A system for compensating gain control variations in a power amplifier comprises a power control element configured to receive a power control signal and an instantaneous envelope power reference signal, an adder configured to combine the power control signal and the instantaneous envelope power reference signal to obtain a modified power level signal, and a mapping function configured to receive the modified power level signal and configured to alter a control input to a variable gain amplifier, the variable gain amplifier controlling an adjustable input to the power amplifier.

Description:
BACKGROUND  
       [0001]     Radio frequency (RF) transmitters are found in many one-way and two-way communication devices, such as portable communication devices (cellular telephones), personal digital assistants (PDAs) and other communication devices. An RF transmitter must transmit using whatever communication methodology is dictated by the particular communication system within which it is operating. For example, communication methodologies typically include amplitude modulation, frequency modulation, phase modulation, or a combination of these. In a typical global system for mobile communications (GSM) communication system using narrowband TDMA technology, a GMSK modulation scheme supplies a low noise phase modulated (PM) transmit signal to a non-linear power amplifier directly from an oscillator.  
         [0002]     In such an arrangement, a non-linear power amplifier, which is highly efficient, can be used, thus allowing efficient transmission of the phase-modulated signal and minimizing power consumption. Because the modulated signal is supplied directly from an oscillator, the need for filtering, either before or after the power amplifier, is minimized. Other transmission standards, such as that employed in IS-136, however, use a modulation scheme in which the transmitted signal is both phase modulated (PM) and amplitude modulated (AM). Standards such as these increase the data rate without increasing the bandwidth of the transmitted signal. Unfortunately, existing GSM transmitter hardware is not easily adapted to transmit a signal that includes both a PM component and an AM component. One reason for this difficulty is that in order to transmit a signal containing a PM component and an AM component, a highly linear power amplifier is required. Unfortunately, highly linear power amplifiers are very inefficient, thus consuming significantly more power than a non-linear power amplifier and drastically reducing the talk-time and standby time of the portable communication device on a battery charge.  
         [0003]     This condition is further complicated because GSM transmitters transmit in bursts and must be able to control the ramp-up of the transmit power as well as have a high degree of control over the output power level over a wide power range. In GSM this power control is typically performed using a closed feedback loop in which a portion of the signal output from the power amplifier is compared with a reference signal and the resulting error signal is fed back to the control port of the power amplifier.  
         [0004]     When attempting to include an AM component in a phase modulated GSM type modulation system, the power control loop will attenuate the amplitude variations present in the signal in an attempt to maintain a constant output power. In such an arrangement, the power control loop tends to cancel the AM portion of the signal.  
         [0005]     In such systems in which transmit signals contain both PM and AM components, the output power can be controlled by applying a pre-determined control voltage to the power amplifier. Unfortunately, this requires the use of a power amplifier with a highly linear control characteristic and wide dynamic control range. In general the highly efficient power amplifiers used in GSM transmitters do not normally exhibit these properties to a sufficient degree. In non-burst transmission systems the output power may be controlled by a feedback loop having a time-constant that is very low compared to the time-constant of the amplitude variations of the modulator. Another known method to control the output power is to “pre-distort” the modulated signal in such a way that the power control loop will cancel the effect of the pre-distortion. In such a method, the amplitude information is passed through a transfer function that is the inverse of the power control loop transfer function. Unfortunately, these methods are costly, inefficient and require significant calibration time, dedicated test equipment sensors and fine control over the input power to the power amplifier.  
         [0006]     Further, in those transmission standards in which the signal sent to a power amplifier contains both a PM and an AM component, unless the power amplifier is very linear, it may distort the combined transmission signal by causing undesirable AM to PM conversion. This conversion is detrimental to the transmit signal and can require the use of a costly and inefficient linear power amplifier.  
         [0007]     Further still, in transmission systems in which a combined AM and PM signal is used in a closed power control loop, it is difficult to obtain the full dynamic range in the AM signal to encompass all output power levels and to obtain sufficient dynamic range to smoothly control the ramp-up and ramp-down of the output power. In a closed power control loop system, it is important to maintain a constant, or nearly constant, loop parameters. In a system in which the AM is injected onto the transmit signal via the AM power control loop, the bandwidth requirements of the power control loop (both absolute value and consistency over the power range) are quite stringent. When a non-linear power amplifier is used to improve overall system efficiency, the gain control characteristics of the power amplifier tend to vary significantly over the range of operation and with temperature change of the power amplifier. This variation of the gain control characteristics impede system performance parameters, such as loop bandwidth, modulation accuracy, loop stability, variation in the power amplifier/system turn-on threshold (which can be thought of as a lowest reliable controlled output power), and other parameters. A significant portion of the variation of the gain control characteristics is due to the gain control function of the power amplifier (PA). Compensating the gain control characteristic of the PA by predistorting the transfer function of the PA gain control element is extremely difficult because the slope of the gain control signal could be as high as 150-250 dB/volt, especially at low power levels, thereby making predistortion of the transfer function of the gain control signal impractical.  
       SUMMARY  
       [0008]     A system for compensating gain control variations in a power amplifier comprises a power control element configured to receive a power control signal and an instantaneous envelope power reference signal, an adder configured to combine the power control signal and the instantaneous envelope power reference signal to obtain a modified power level signal, and a mapping function configured to receive the modified power level signal and configured to alter a control input to a variable gain amplifier, the variable gain amplifier controlling an adjustable input to the power amplifier.  
         [0009]     Related methods of operation are also provided. Other systems, methods, features, and advantages of the invention will be or become apparent to one with skill in the art upon examination of the following figures and detailed description. It is intended that all such additional systems, methods, features, and advantages be included within this description, be within the scope of the invention, and be protected by the accompanying claims. 
     
    
     BRIEF DESCRIPTION OF THE FIGURES  
       [0010]     The invention can be better understood with reference to the following figures. The components within the figures are not necessarily to scale, emphasis instead being placed upon clearly illustrating the principles of the invention. Moreover, in the figures, like reference numerals designate corresponding parts throughout the different views.  
         [0011]      FIG. 1  is a block diagram illustrating a simplified portable transceiver.  
         [0012]      FIG. 2  is a block diagram illustrating the system and method for power mapping of the invention.  
         [0013]      FIG. 3  is a graphical representation of the power output of power amplifier of  FIG. 2  versus the voltage V PC  applied to the control input of the power amplifier.  
         [0014]      FIG. 4  is a graphical representation of the power amplifier gain control characteristic and the characteristic of the power mapping function of  FIG. 2 .  
         [0015]      FIG. 5  is a graphical representation of the power amplifier gain versus the output, P OUT,dBm , of the power amplifier.  
         [0016]      FIG. 6  is a flow chart illustrating the operation of an embodiment of the system and method for power mapping. 
     
    
     DETAILED DESCRIPTION  
       [0017]     Although described with particular reference to a portable transceiver, the system and method for power mapping to compensate power amplifier gain control variations, referred to as the system and method for power mapping, can be implemented in any system where it is desirable to transmit a combined signal including a PM component and an AM component and maintain high dynamic range control over the AM signal.  
         [0018]     The system and method for power mapping can be implemented in hardware, software, or a combination of hardware and software. When implemented in hardware, the system and method for power mapping can be implemented using specialized hardware elements and logic. When the system and method for power mapping is implemented partially in software, the software portion can be used to precisely control the mapping function in the power control loop. The software can be stored in a memory and executed by a suitable instruction execution system (microprocessor). The hardware implementation of the system and method for power mapping can include any or a combination of the following technologies, which are all well known in the art: discrete electronic components, a discrete logic circuit(s) having logic gates for implementing logic functions upon data signals, an application specific integrated circuit having appropriate logic gates, a programmable gate array(s) (PGA), a field programmable gate array (FPGA), etc.  
         [0019]     The software for the system and method for power mapping comprises an ordered listing of executable instructions for implementing logical functions, and can be embodied in any computer-readable medium for use by or in connection with an instruction execution system, apparatus, or device, such as a computer-based system, processor-containing system, or other system that can fetch the instructions from the instruction execution system, apparatus, or device and execute the instructions.  
         [0020]     In the context of this document, a “computer-readable medium” can be any means that can contain, store, communicate, propagate, or transport the program for use by or in connection with the instruction execution system, apparatus, or device. The computer readable medium can be, for example but not limited to, an electronic, magnetic, optical, electromagnetic, infrared, or semiconductor system, apparatus, device, or propagation medium. More specific examples (a non-exhaustive list) of the computer-readable medium would include the following: an electrical connection (electronic) having one or more wires, a portable computer diskette (magnetic), a random access memory (RAM), a read-only memory (ROM), an erasable programmable read-only memory (EPROM or Flash memory) (magnetic), an optical fiber (optical), and a portable compact disc read-only memory (CDROM) (optical). Note that the computer-readable medium could even be paper or another suitable medium upon which the program is printed, as the program can be electronically captured, via for instance, optical scanning of the paper or other medium, then compiled, interpreted or otherwise processed in a suitable manner if necessary, and then stored in a computer memory.  
         [0021]      FIG. 1  is a block diagram illustrating a simplified portable transceiver  100 . The portable transceiver  100  includes speaker  102 , display  104 , keyboard  106 , and microphone  108 , all connected to baseband subsystem  110 . In a particular embodiment, the portable transceiver  100  can be, for example but not limited to, a portable telecommunication device such as a mobile cellular-type telephone. Speaker  102  and display  104  receive signals from the baseband subsystem  110  via connections  112  and  114 , respectively, as known to those skilled in the art. Similarly, keyboard  106  and microphone  108  supply signals to the baseband subsystem  110  via connections  116  and  118 , respectively. The baseband subsystem  110  includes microprocessor (μP)  120 , memory  122 , analog circuitry  124 , and digital signal processor (DSP)  126  in communication via bus  128 . The bus  128 , though shown as a single bus, may be implemented using a number of busses connected as necessary among the subsystems within baseband subsystem  110 . The microprocessor  120  and the memory  122  provide the signal timing, processing and storage functions for the portable transceiver  100 . If portions of the system and method for power mapping are implemented in software, then the memory  122  also includes power mapping software  255  that can be executed by the microprocessor  120 , DSP  126  or by another processor.  
         [0022]     The analog circuitry  124  provides the analog processing functions for the signals within baseband subsystem  110 . The baseband subsystem  110  provides control signals to radio frequency (RF) subsystem  130  via connection  132 . In one embodiment, the baseband subsystem  110  provides an automatic power control signal (supplied as an analog voltage signal and referred to as V APC ) to the RF subsystem  130 . Although shown as a single connection  132 , the control signals may originate from the DSP  126  from microprocessor  120 , or from another element, and are supplied to a variety of points within the RF subsystem  130 . It should be noted that, for simplicity, only the basic components of portable transceiver  100  are illustrated.  
         [0023]     The baseband subsystem  110  also includes analog-to-digital converter (ADC)  134  and digital-to-analog converters (DACs)  136  and  142 . The ADC  134  and DACs  136  and  142  also communicate with microprocessor  120 , memory  122 , analog circuitry  124  and DSP  126  via bus  128 . While shown as two separate DACs  136  and  142 , a single DAC may be implemented. The DAC  136  converts the digital communication information within baseband subsystem  110  into an analog signal for transmission to RF subsystem  130  via connection  140 . Connection  140 , while shown as two directed arrows, includes the information that is to be transmitted by RF subsystem  130  after conversion from the digital domain to the analog domain. The DAC  136  supplies baseband in-phase (I) and quadrature-phase (Q) components of the information signal to be transmitted via connection  140  to the modulator  146 . In such an embodiment, modulator  146  is an I/Q modulator as known in the art. Other modulation methodologies may also be employed. The DAC  142  supplies control signals to various components with RF subsystem  130  via connection  132 .  
         [0024]     The modulator  146  modulates the I and Q information signals received from the DAC  136  onto an LO signal and provides a modulated signal via connection  152  to upconverter  154 . The modulator  146  also supplies an intermediate frequency (F) signal containing only the desired amplitude modulated (AM) signal component on connection  138  for input to the power control element  300  via connection  138 . In an embodiment, as will be described below, the AM signal supplied by the modulator via connection  138  can first be supplied to a reference variable gain element, and then supplied to a detector and a logarithmic amplifier associated with the RF subsystem  130 ,. The AM signal supplied by the modulator  146  is an intermediate frequency (IF) AM signal with a constant (average) power level that is supplied as a reference signal (V REF ) to the reference variable gain element to be described below. The power control element  300  also supplies to the upconverter  154  via connection  144  a constant level IF signal containing the phase modulated (PM) component of the transmit signal. The operation of the power control element  300  will be described below with reference to  FIG. 2 .  
         [0025]     The upconverter  154  receives a frequency reference signal (referred to as a “local oscillator” or “LO” signal) from synthesizer  148  via connection  156 . The synthesizer  148  determines the appropriate frequency to which the upconverter  154  will translate the modulated signal on connection  152 .  
         [0026]     The upconverter  154  supplies a phase modulated signal at the appropriate transmit frequency via connection  158  to power amplifier  160 . The power amplifier  160  amplifies the phase-modulated signal on connection  158  to the appropriate power level and applies the amplitude modulation for transmission via connection  162  to antenna  164 . Illustratively, switch  166  controls whether the amplified signal on connection  162  is transferred to antenna  164  or whether a received signal from antenna  164  is supplied to filter  168 . The operation of switch  166  is controlled by a control signal from baseband subsystem  110  via connection  132 .  
         [0027]     A portion of the amplified transmit signal power on connection  162  is supplied via connection  170  to power control element  300 . The power control element  300  forms a continuous closed power control feedback loop and supplies an information signal, referred to as V PC , on connection  172  instructing the power amplifier  160  as to the power to which the signal on connection  158  should be amplified. The power control element  300  also receives the LO signal from synthesizer  148  via connection  198 . The operation of power control element  300  will be described in further detail with respect to  FIG. 2 .  
         [0028]     A signal received by antenna  164  may, at the appropriate time determined by baseband subsystem  110 , be directed via switch  166  to a receive filter  168 . The receive filter  168  filters the received signal and supplies the filtered signal on connection  174  to a low noise amplifier (LNA)  176 . The receive filter  168  may be a bandpass filter that passes all channels of the particular cellular system where the portable transceiver  100  is operating. As an example, for a 900 MHz GSM system, receive filter  168  would pass all frequencies from 935.1 MHz to 959.9 MHz, covering all 124 contiguous channels of 200 kHz each. The purpose of the receive filter  168  is to reject all frequencies outside the desired region. An LNA  176  amplifies the very weak signal on connection  174  to a level at which downconverter  178  can translate the signal from the transmitted frequency back to a baseband frequency. Alternatively, the functionality of the LNA  176  and the downconverter  178  can be accomplished using other elements, such as, for example but not limited to, a low noise block downconverter (LNB).  
         [0029]     The downconverter  178  receives an LO signal from synthesizer  148  via connection  180 . The LO signal determines the frequency to which to downconvert the signal received from the LNA  176  via connection  182 . The downconverted frequency is called the intermediate frequency (IF). The downconverter  178  sends the downconverted signal via connection  184  to a channel filter  186 , also called the “IF filter.” The channel filter  186  filters the downconverted signal and supplies it via connection  188  to an amplifier  190 . The channel filter  186  selects the one desired channel and rejects all others. Using the GSM system as an example, only one of the  124  contiguous channels is actually to be received. After all channels are passed by the receive filter  168  and downconverted in frequency by the downconverter  178 , only the one desired channel will appear precisely at the center frequency of channel filter  186 . The synthesizer  148 , by controlling the local oscillator frequency supplied on connection  180  to downconverter  178 , determines the selected channel. The amplifier  190  amplifies the received signal and supplies the amplified signal via connection  192  to demodulator  194 . The demodulator  194  recovers the transmitted analog information and supplies a signal representing this information via connection  196  to the ADC  134 . The ADC  134  converts these analog signals to a digital signal at baseband frequency and transfers them via bus  128  to DSP  126  for further processing.  
         [0030]      FIG. 2  is a block diagram  200  illustrating an embodiment of the system and method for power mapping. Beginning with a description of the power control loop  300 , a portion of the output power present at the output of power amplifier  160  on connection  162  is diverted by a coupler  222  via connection  170  and input to a mixer  226  in the feedback path. The mixer  226  also receives the local oscillator (LO) signal from synthesizer  148  via connection  198 .  
         [0031]     The mixer  226  downconverts the RF signal on connection  170  to an intermediate frequency (IF) signal on connection  228 . For example, the mixer  226  translates a signal having a frequency of approximately 2 gigahertz (GHz) on connection  170  to a frequency of approximately 100 megahertz (MHz) on connection  228  for input to the variable gain element  232 . The variable gain element  232  can be, for example but not limited to, a variable gain amplifier or an attenuator. In such an arrangement, the variable gain element  232  might have a total variable gain range of approximately 70 dB. The variable gain element  232  receives a control signal input from the inverting output of amplifier  236  via connection  234 . The input to amplifier  236  is supplied via connection  132  from the DAC  142  of  FIG. 1 . The signal on connection  132  is the reference voltage automatic power control signal, V APC , that determines the transmit power level and provides the power profile. The V APC  signal on connection  132  is supplied to a reconstruction filter, which includes resistor  240  and capacitor  242 . In this manner, a power control signal for the transmit power level and power profile is supplied via connection  234  to the control input of the variable gain element  232 .  
         [0032]     The output of variable gain element  232  on connection  246  is at an IF and includes modulation having both an AM component and a PM component and is called the “power measurement signal.” This power measurement signal is related to the absolute output power of power amplifier  160 , and includes a very small error related to the AM and PM components present in the signal. The output of variable gain element  232  on connection  246  is supplied to the input of a power detector  262  and is also supplied to a limiter  248  in the phase locked loop  220 , although the limiter  248  may reside outside of the PLL  220 .  
         [0033]     The IF signal on connection  246  includes both an AM component and a PM component. The IF signal on connection  246  is supplied to the power detector  262 , which provides, on connection  264 , a baseband signal representing the instantaneous level of IF power present on connection  246 . The output of power detector  262  on connection  264  is supplied to the inverting input of amplifier  268 .  
         [0034]     The amplifier  268 , capacitor  266  and capacitor  270  form a comparator  284 , which provides the error signal used to control the power amplifier  160  via connection  272 . The non-inverting input to the amplifier  268  is supplied via connection  138  from a reference power detector  276 . The output of the modulator  146  on connection  252  is supplied to a reference variable gain element  250 . The reference variable gain element  250  can be implemented as a variable gain amplifier (VGA), which is similar to the variable gain elements  232  and  274 , but which has different gain characteristics. The signal supplied to the reference VGA on connection  252  comprises an IF signal containing the amplitude modulation (AM) portion of the modulated signal, which is an instantaneous envelope reference signal. The reference VGA  250  receives the V APC  signal via connection  132  to its control input. The reference VGA  250  extends the dynamic range of the AM control signal that is supplied through the reference power detector  276  to the non-inverting input of the amplifier  268  in the comparator  284 . The signal on connection  138  supplied to the non-inverting input of amplifier  268  contains the demodulated amplitude information of the AM signal developed by the modulator  146  in the phase locked loop  220 .  
         [0035]     The gain of the power control loop  300  amplifies the signal on connection  272  such that the difference between the signals on connections  264  and  138  input to amplifier  268  provide an error signal on connection  272  that is used to control the output of the power amplifier  160 . In accordance with an embodiment of the invention, the output of the power detector  276  is also supplied to a logarithmic (log) amplifier  290 . The log amplifier  290  converts the instantaneous envelope reference signal, V REF , on connection  138  to the logarithmic domain on connection  291 . The signal on connection  291  is referred to as a compensation signal, V COMP . Alternatively, a logarithmic detector (not shown) can be used to take the signal on connection  258  and create the signal on connection  291 . If a logarithmic detector is implemented, the power detector  276  will still be used to supply the reference signal to the power control element  300 . The signal on connection  291  is supplied to an adder  298 . The adder  298  combines the log reference signal, V COMP , on connection  291  and the V APC  signal on connection  133 . The output of the adder  298  is supplied to a mapping function  296 . The mapping function  296  generates a control signal that is supplied to a variable gain amplifier  275 . By combining the log reference signal, V COMP , and the V APC  signal, gain control variations of the power amplifier  160  can be compensated by the variable gain amplifier  275  using both an average power level (the V APC  signal) and the instantaneous envelope (the V REF  signal, corresponding to the AM signal acted upon by the log amplifier  290 ). In accordance with an embodiment of the invention, the mapping function  296  takes the output of the adder  298  and alters the gain of the variable gain element  275  to compensate for gain control variations of the power amplifier  160 , independent of temperature variations of the power amplifier  160 . By knowing the value of the power control signal V APC  and the logarithmic representation of the instantaneous envelope reference signal V REF , the transfer function of the power control loop  300  can be linearized, independent of temperature. Further, the bandwidth of the power control loop remains constant over the entire power control range.  
         [0036]     The error signal on connection  272  is supplied to a low pass filter  292 . The output of the low pass filter  292  on connection  294  is supplied to the variable gain element  275 . The output of the variable gain element  275 , which is a signal that compensates for the gain control variations of the power amplifier  160 , is supplied to the variable gain amplifier  274 , which can be similar in structure to the variable gain element  232 . However, the variable gain element  274  has a gain function that is the inverse of the gain function of the variable gain element  232 , since the control input to the variable gain element  274  is supplied from the non-inverting output of amplifier  236  and the control input to the variable gain element  232  is supplied from the inverting output of amplifier  236 . The control input to the variable gain amplifier  275  is controlled by the power mapping function  296  to compensate for gain variations in the control input to the power amplifier  160 . In an alternative embodiment, the functions of the variable gain amplifiers  274  and  275  can be combined into a single variable gain amplifier. In this manner, the power amplifier control signal on connection  172  supplied to the control port of power amplifier  160  drives the power amplifier  160  to provide the proper output on connection  162 , while compensating for gain control variations of the power amplifier  160 .  
         [0037]     The level of the signal on connection  264  and the level of the signal on connection  138  will be nearly equal, by the function of the negative feedback loop in the power control element  300 . For example, if the gain of the variable gain element  232  is increased by a factor of 10, then the level of the output of power amplifier  160  will decrease accordingly to maintain equilibrium at the input of the amplifier  268 . The output of the power amplifier  160  changes to cancel the gain change of the variable gain element  232 . In this manner, the feedback amplitude signal on connection  264  remains nearly equal to the reference amplitude signal on connection  138 . In this manner, a feedback signal containing an AM and a PM portion is present on connection  246 . The signal on connection  246  is converted by power detector  262  from an IF signal to a baseband signal on connection  264 . The difference between the signal on connection  264  and the signal on connection  138  is amplified by amplifier  268  and amplifier  274  to drive the power amplifier control port on connection  172  so that the desired signal is achieved at the output of the power amplifier  160  on connection  162 . The power control loop  300  has sufficient gain so that the error signal on connection  272  can be kept small. In such a case, the gain changes of the variable gain element  232  and the power amplifier  160  will substantially be the inverse of each other.  
         [0038]     The amplifier  268  compares the power measurement signal on connection  264  with a reference voltage signal from the reference power detector  276  on connection  138 , representing the AM portion of the signal supplied by the modulator  146 . The output of amplifier  268  on connection  272  is the amplitude error signal. The DC voltage level on connection  138  affects the desired static output power for the power amplifier  268 , irrespective of AM modulation. The amplifier  268  compares the signal level on connection  264  with the signal level on connection  138  and then amplifies the difference, thus providing an error signal on connection  272 . The comparator  284  functions as an integrator, which is also a low pass filter.  
         [0039]     The reference VGA  250  extends the amplitude control range of the power control element  300 . The reference VGA  250 , which is located outside of the amplitude control loop, controls the level of the reference signal supplied to the power control element  300 . The gain characteristic of the reference VGA  250  differs from the gain characteristic of the variable gain elements  232 ,  274  and  275 . In one embodiment, the reference VGA  250  maintains a constant gain over the majority of the amplitude range of the power control element  300 , including the highest output amplitude levels. However, when the output amplitude is at the low end of the range (the range of the analog power control signal V APC ), i.e., when the value of V APC  is low, the gain of the reference VGA  250  is reduced. In this manner, the reference VGA  250  maintains closed loop amplitude control down to the lowest output power levels, and particularly, during transmit power ramp-up and transmit power ramp-down, when V APC  is at a predetermined minimum value. The gain range of the reference VGA  250  is realized as what is referred to as a “soft step” function, and is not linear. The gain of the reference VGA  250  is shifted down by an appropriate amount (for example 15 or 18 dB) when the power control voltage V APC  is a level sufficiently low to indicate that the transmitter (e.g., the upconverter  154  and the power amplifier  160 ) is in either ramp-up or ramp-down and not at a constant output power level. The dynamic range of the power detectors  262  and  276  limits the amplitude control provided by the variable gain element  232  and the variable gain element  274 . The reference VGA  250  provides amplitude control at power output levels lower than the minimum power output associated with a minimum V APC  signal level. For example, this “soft step” function can be implemented for a V APC  signal level of approximately 0.6 V when it is known that the power control characteristic of the power amplifier  160  will always generate output power levels below 0 dBm for this value of V APC  voltage.  
         [0040]     The power amplifier control signal on connection  172  is driven by the variable gain element  274 , which corrects for the effect that variable gain element  232  and the control input to the power amplifier  160  have on the transfer function of the power control loop  300 . The variable gains of variable gain element  232  and variable gain element  274  are complimentary, while the variable gain element  275  compensates for gain variations in the power amplifier  160 . Because the feedback power measurement signal is present on connection  264  and the amplitude reference signal is present on connection  138 , the amplifier  268  provides a dual function; (1) it amplifies the AM error signal (the difference between the signal on connection  138  and the signal on connection  264 ) so as to modulate the power output of power amplifier  160  via connection  172  to have the correct amount of AM; and (2) it performs the average power comparison and amplifies the result, thus providing a control signal also via connection  172  that drives the power amplifier  160  to the correct average power output. Therefore, at connection  172  both the AM error signal and the power control error signal are amplified to a level sufficient to drive the power amplifier  160  to the desired average power with the desired AM signal. In this manner, the desired AM portion of the signal is supplied to the control input  172  of power amplifier  160  and made present on the power amplifier output on connection  162 . The mixer  226 , variable gain element  232 , power detector  262 , amplifier  268  and variable gain elements  274  and  275  provide a continuous closed-loop power control feedback system to control the power output of power amplifier  160 , while allowing for the introduction of the AM portion of the transmit signal via connection  138 .  
         [0041]     At all times, the continuous power-control feedback loop allows the correction of any phase shift caused by power amplifier  160 . In this manner, the PLL  220  includes a feedback loop for looping back the output of power amplifier  160  to the input of phase/frequency detector  208 . Any unwanted phase shift generated by the power amplifier  160  will be corrected by the PLL  220 . The output of variable gain element  232  passes any phase distortion present via connection  246  to limiter  248  for correction by the PLL  220 . As such, the phase of the output of power amplifier  160  is forced to follow the phase of the LO signal on connection  156 .  
         [0042]     In order to remove the AM from the output of variable gain element  232 , the variable gain element  232  is connected via connection  246  and connection  144  to the input of limiter  248 . The limiter  248  develops a feedback signal containing only a PM component on connection  206 . The baseband I and Q information signals are supplied to the modulator  146  via connections  278  and  282 , respectively. The I and Q baseband information signal interface is understood by those skilled in the art. As a result of the operation of the modulator  146 , the output on connection  252  is an intermediate frequency signal including an AM component providing an AM reference signal and a PM component providing a PM reference signal. The output of modulator  146  is supplied via connection  252  to the reference VGA  250 . The output of the reference VGA  250  is supplied to the reference power detector  276  to ensure that the reference power detector  276  receives a signal having constant average power, thus reducing the dynamic range requirement of the reference power detector  276  to only what is needed to cover the modulation range, which is typically less than 20 dB. The output of the reference power detector  276  is a voltage signal representing the amplitude of the desired transmit signal. In this manner, the signal provided on connection  138  is a reference signal that allows amplitude control of the power control loop  300  at power output levels over the full output power range, thus expanding the closed loop power control range of the power control loop  300  to over 65 dB.  
         [0043]     The modulator  146  also provides the PM component of the signal on connection  252 . This PM signal is then supplied to limiter  249 , which outputs a signal on connection  256  containing the phase reference component of the transmit signal. The components within the phase locked loop  220  provide gain for the comparison of the PM on connection  256  and the phase feedback signal on connection  206 , thus providing a phase error output of the phase detector  208  on connection  210 . In this manner, a feedback signal taken from the output of variable gain element  232  on connection  246  is supplied as continuous feedback to the phase locked loop  220 .  
         [0044]     It should be noted that even when the power amplifier  160  is not operating, there will always be some small leakage through the power amplifier  160  onto connection  162 . This small leakage is sufficient to provide a feedback signal through the variable gain element  232  and into the phase locked loop  220  such that the phase locked loop  220  can be locked using just the leakage output of power amplifier  160 . In this manner, a single feedback loop can be used to continuously control the output power of power amplifier  160  from the time that the amplifier is off through the time when the amplifier  160  is providing full output power.  
         [0045]     The modulator  146  receives an LO input signal from synthesizer  148  via connection  156 . The LO signal is frequency divided by a number “x” in order to provide a signal having an appropriate frequency on connection  204 . The number “x” is chosen so as to minimize the design complexity of the synthesizer  148  and can be, for example, but not limited to, chosen to convert the output of the synthesizer  148  to a frequency of about 100 MHz.  
         [0046]     The output of phase/frequency detector  208  on connection  210  is a digital signal having a value of either a 0 or a 1 with a very small transition time between the two output states. This signal on connection  210  is supplied to low-pass filter  212 , which integrates the signal on connection  210  and places a DC signal on connection  214  that controls the frequency of the transmit voltage control oscillator (TX VCO)  216 . The output of TX VCO  216  is supplied via connection  158  directly to the power amplifier  160 . In this manner, the synthesizer  148 , limiter  248 , modulator  146 , limiter  249 , divider  260 , divider  202 , phase/frequency detector  208 , low-pass filter  212  and TX VCO  216  form a phase locked loop (PLL)  220 , which is used to determine the transmit frequency on connection  158 . When the PLL  220  is settled, or “locked,” then the two signals entering the phase/frequency detector  208  on connections  256  and  206  have precisely the same phase and frequency, and the output of the phase/frequency detector  208  on connection  210  goes to zero. The output of the integrating low-pass filter  212  on connection  214  stabilizes, resulting in a fixed frequency out of TX VCO  216 . For example, the synthesizer  148  and the mixer  226  ensure that the frequency of the signal output from the TX VCO  216  on connection  158  tracks the sum of the frequencies of the local oscillator signal supplied by synthesizer  148  and the IF frequency on connection  206 .  
         [0047]     When the phase locked loop  220  is locked, the phase of the signal on connection  256  and the phase of the signal on connection  206  will be equal. The gain in the phase locked loop  220  has to be sufficiently high to amplify the error signal on connection  206  to a level at which the phase/frequency detector  208  can make a comparison. By using the modulator  146  to impose the I and Q information signals on the signal on connection  204 , and because the phase locked loop  220  will keep the phase of the TX VCO locked, the phase of the signal output from the TX VCO  216  on connection  158  will follow that of the phase imposed by the modulator  146 . In this manner, the PM error signal present on connection  210  is minimized by the very high sensitivity, of the order of many MHz per volt, of the TX VCO  216 .  
         [0048]     Because the power control loop  300  is a closed loop for AM signals at connection  138 , it is possible to use a non-linear, and therefore highly efficient, power amplifier  160 . Furthermore, the undesirable and detrimental AM-to-PM conversion, which occurs due to the amplitude dependence of an amplifier&#39;s phase shift, is rectified by the power amplifier  160  being included within the phase locked loop  220 . By separating the AM and the PM modulation and by providing closed loop control for both the AM and PM modulation, a non-linear, and therefore highly efficient power amplifier can be used.  
         [0049]     In some applications it is desirable to allow the power amplifier  160  to output a signal containing both an AM component and a PM component, while maintaining the power amplifier  160  as a non-linear (and therefore highly efficient) power amplifier. In such a case, the output of modulator  146  will include both an AM and a PM component, with limiter  249  used to cancel the AM component present on connection  252 , thereby preventing any AM-to-PM conversion in the phase/frequency detector  208 .  
         [0050]      FIG. 3A  is a graphical representation  350  of the power amplifier gain control as a function of the power amplifier control voltage V PC . The horizontal axis  352  of the graph  350  represents the voltage V PC  applied to the control input of the power amplifier  160  and the vertical axis  354  represents the control gain of the power amplifier  160 . When the control gain of the power amplifier is G=Δ V RFOUT /Δ V PC , with respect to control voltage of the power amplifier, the power amplifier control gain varies substantially over temperature as illustrated using gain trace  356  at temperature t 1  and gain trace  357  at temperature t 2 . For a given control voltage, V PC , the slope of the power amplifier gain control can vary from several dB/V to approximately 150-200 dB/V over a relatively narrow voltage range, as illustrated using reference numeral  358 . This variation over a relatively narrow voltage range makes it very difficult to accurately compensate the gain variation over a number of different parts and over a temperature range without the need for extensive and complicated calibration procedures.  
         [0051]      FIG. 3B  is a graphical representation  370  of the power amplifier gain control as a function of the power output of the power amplifier. The horizontal axis  372  of the graph  350  represents the power output, P OUT,dBm , of the power amplifier  160  and the vertical axis  374  represents the control gain of the power amplifier  160 , G=(Δ V RFOUT /ΔV PC ). When the control gain of the power amplifier (G=Δ V RFOUT /Δ V PC ) is varied with respect to output power of the power amplifier, the power amplifier control gain varies substantially less over temperature than when the gain control of the power amplifier is varied as a function of the power amplifier control voltage. Furthermore, the power amplifier gain control is relatively stable over temperature and from part to part. As shown in  FIG. 3B , the gain trace  376  at temperature t 1  and the gain trace  377  at temperature t 2  vary little over temperature when the gain control of the power amplifier is drawn as a function of the output power of the power amplifier, and as shown using reference numeral  378 . Therefore, if the gain variation of the power amplifier is compensated as a function of the output power (P OUT,dBm ), variation from part to part and variation over temperature are significantly reduced when compared to the case when power amplifier gain compensation is performed as a function of its control input voltage (V PC ).  
         [0052]     In this embodiment, the reference information is the output power of the power amplifier, instead of the power control signal, V PC . Thus, the gain compensation (to maintain the same loop gain-bandwidth) is performed in the closed power control loop. In this manner, the power amplifier gain control is based on the actual power amplifier output power, and not on the power amplifier control voltage. Therefore, the power amplifier gain compensation mapping function is basically a complimentary curve of the power amplifier gain control as a function of the output power. The power amplifier output power is a combination of the detected envelope (instantaneous variations) and the operating power set by the power control signal, as shown below in  FIG. 4 .  
         [0053]      FIG. 4  is a graphical representation  400  of the power amplifier gain control characteristic and the characteristic of the power mapping function of  FIG. 2 . The horizontal axis  402  represents the output, P OUT,dBm , of the power amplifier and the vertical axis  404  represents the value of the gain in dB. The curve  406  represents 0 dB gain and the curve  408  represents the total gain of the power control loop, in this example, 20 dB. The curve  410  represents the gain applied to the variable gain amplifier  275 , as controlled by the power mapping function  296 , and the curve  412  represents the control gain of the power amplifier  160 . The gain applied to the variable gain amplifier  275 , as controlled by the power mapping function  296 , is the reciprocal function of the gain applied to the control input of the power amplifier  160 . In this manner, any gain variations in the control input to the power amplifier  160  are compensated by the power mapping function  296 .  
         [0054]      FIG. 5  is a graphical representation  500  of the power amplifier gain versus the output, P OUT,dBm , of the power amplifier. The horizontal axis  502  represents the output, P OUT,dBm , of the power amplifier and the vertical axis  404  represents the value of the gain in dB. The curve  505  illustrates the gain of the control input of the power amplifier  160 . The power control signal, V APC , is shown as a window  506 , and the instantaneous envelope reference signal, V REF , is shown at the center of the window  506 . The point  510  represents the point at which the transfer function of the power control loop exactly compensates for gain control variations in the power amplifier  160 . The portion  515  of the curve  505  represents the range of compensation available for a given value of V APC .  
         [0055]      FIG. 6  is a flow chart  600  illustrating the operation of an embodiment of the system and method for power mapping. The blocks in the flow chart can be executed in the order shown, out of the order shown, or substantially in parallel. In block  602 , the V APC  signal is supplied the power control element  300 . In block  604 , an instantaneous envelope reference level signal, V REF , is supplied to the power control element  300 . In block  606 , the instantaneous envelope reference level signal, V REF , is converted to the logarithmic domain as signal V COMP . In an alternative embodiment, a logarithmic detector can be substituted for the power detector  276  and the log amplifier  290 , so that a logarithmic signal representing the instantaneous envelope reference level signal, V REF , is available to be supplied to the power control element  300 . In block  608 , the logarithmic representation, V COMP , of the instantaneous envelope reference level signal, V REF , and the power control signal, V APC , are combined. In block  610 , a power mapping function is applied to the resultant signal. In block  612 , the power amplifier control signal, VPC, is supplied to the power amplifier  160 . The signal V PC  represents a linearized control signal provided by the power control loop  300 .  
         [0056]     While various embodiments of the invention have been described, it will be apparent to those of ordinary skill in the art that many more embodiments and implementations are possible that are within the scope of this invention. Accordingly, the invention is not to be restricted except in light of the attached claims and their equivalents.