Abstract:
A method and apparatus for estimating mobile radio channel parameters by dynamically determining the delay offset, carrier frequency offset, and coherent averaging length that yields the optimum despreading gain for current channel conditions. Obtaining the optimal despreading gain increases the fidelity of desired channel parameter estimates, (such as delay, phase, and the complex impulse response), and also expands the range of conditions under which they can be feasibly measured. A plurality of delay offsets, carrier frequency offsets, and coherent averaging lengths are considered for each measurement of channel parameters. An energy metric assigned to each combination of delay offset and coherent averaging length, and this value measures the despreading gain for these conditions.

Description:
FIELD OF THE INVENTION 
     The present invention relates generally to communication systems and, in particular, to a method and apparatus for estimating one or more radio channel parameters within a mobile communication system. 
     BACKGROUND OF THE INVENTION 
     Radio location techniques are widely used in many mobile transmitter systems. Among these location techniques are a) Time of Arrival (TOA), b) Time Difference of Arrival (TDOA), and c) Angle of Arrival (AOA) methods. Each of these techniques requires either the measurement or estimation of one or more parameters of the communication channel utilized by the mobile transmitter. 
     To estimate the location of a mobile transmitter in a code division multiple access (CDMA) communication system, estimates of channel parameters such as delay, amplitude, and possibly phase must be obtained for multiple signals. In general, these parameter estimates will be obtained by correlating received signals with one or more “perfect” reference signals. 
     Correlation with a reference waveform can also be interpreted as despreading followed by coherent averaging. Up to a point, the longer the correlation (coherent averaging length), the more despreading gain is obtained. Effects from noise and interference on these channel estimates, which are normally severe, can be made arbitrarily small so long as the despreading gain can be increased. In practice, the maximum despreading gain is limited by the non-stationarity of the communication channel. If the communication channel lacks sufficient stationarity, a longer averaging window will ultimately result in a decrease in despreading gain below the maximum achievable level. Thus, it is desirable to determine the optimal coherent averaging length and the corresponding maximum despreading gain for the radio channel conditions achievable at any given observation. 
     Higher gain is advantageous particularly in mobile transmitter location applications. This is because it may be necessary for an adjacent sector or remote sector to detect the mobile transmitter to accurately fix its position. In a communication system, such as a CDMA communication system where the mobile transmitter is power controlled, when the mobile transmitter is located near to a serving sector, it may communicate with relatively low power. As a result, detection of the mobile transmitter&#39;s signal at adjacent or remote sectors is difficult. Where detection is required, such as when determining the mobile transmitter&#39;s location, it has been proposed to have the mobile transmitter “power-up” to a level sufficient to permit detection at the adjacent or remote serving sector. Such a strategy, however, defeats the benefits of providing power control. 
     Therefore, a need exists for a method and apparatus for channel parameter estimation having high gain under various communication channel conditions. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram of a communication system in accordance with the preferred embodiments of the present invention. 
     FIG. 2 is a block diagram of a first base station of the communication system depicted in FIG.  1  and in accordance with the preferred embodiments of the present invention. 
     FIG. 3 is a block diagram of a second base station of the communication system depicted in FIG.  1  and in accordance with the preferred embodiments of the present invention. 
     FIG. 4 is a block diagram of a frequency domain searcher in accordance with the preferred embodiments of the present invention. 
     FIG. 5 is a graph illustrating a hypothetical power spectrum before decimation within the frequency domain searcher illustrated in FIG.  4 . 
     FIG. 6 is a graph illustrating a hypothetical power spectrum after decimation within the frequency domain searcher illustrated in FIG.  4 . 
     FIG. 7 is graphical representation of zero-padding utilized by the frequency domain searcher illustrated in FIG.  4 . 
     FIG. 8 is a graph illustrating an energy metric calculation in accordance with a preferred embodiment of the invention. 
     FIG. 9 is a graph illustrating an energy metric calculation in accordance with an alternate preferred embodiment of the invention. 
     FIG. 10 is a graph illustrating an averaging window in accordance with a preferred embodiment of the invention. 
     FIG. 11 is a graph illustrating a maximum energy-to-noise ratio calculation in accordance with a preferred embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     To address the above-mentioned need, a method and apparatus for estimating an energy metric is provided herein. In the preferred embodiment, the invention searches through a plurality of delay offsets, carrier frequency offsets, and coherent averaging lengths so as to maximize this metric. Finding the peak value for the energy metric in turn determines the delay offset, carrier frequency offset, and coherent averaging length that yield maximum possible despreading gain for each measurement of channel parameters. 
     With reference now to the drawings wherein like reference numeral are utilized to designate like elements throughout, FIG. 1 is a block diagram of a communication system  100  in accordance with a preferred embodiment of the present invention. In the preferred embodiment, the communication system  100  utilizes a code division multiple access (CDMA) system protocol as described in Cellular System Remote unit-Base Station Compatibility Standard of the Electronics Industry Association/Telecommunications Industry Association Interim Standard 95C (IS-95). (EIA/TIA can be contacted at 2001 Pennsylvania Ave. NW, Washington, D.C. 20006). However, in alternate embodiments, the communication system  100  may utilize other digital cellular communication system protocols such as, but not limited to, the next generation CDMA architecture as described in the UMTS Wideband CDMA SMG2 UMTS Physical Layer Expert Group Tdoc SMG2 UMTS-L1 221/98 (UMTS 221/98), the next generation CDMA architecture as described in the CDMA2000 International Telecommunications Union-Radiocommunication (ITU-R) Radio Transmission Technology (RTT) Candidate Submission document, or the next generation Global System for Mobile Communications (GSM) protocol, the CDMA system protocol as described in “Personal Station-Base Station Compatibility Requirements for 1.8 to 2.0 GHz Code Division Multiple Access (CDMA) Personal Communication Systems” (American National Standards Institute (ANSI) J-STD-008), or the European Telecommunications Standards Institute (ETSI) Wideband CDMA (W-CDMA) protocol. 
     The communication system  100  includes a number of network elements such as a base station  101 , a Centralized Base Station Controller (CBSC)  103 , and a Mobile Switching Center (MSC)  104 . Suitable network elements are commercially available from Motorola, Inc. (Motorola, Inc. is located at 1301 East Algonquin Road, Schaumburg, Ill. 60196). A remote unit  113  is configured to operate within the system  100 . It is contemplated that the network elements within the communication system  100  are configured in well known manners with processors, memories, instruction sets, and the like, which function in any suitable manner to perform the functions set forth herein. 
     As shown, the remote unit  113  is communicating with the base station  101  and the base station  102  via uplink communication signals  119 . The base station  101  is communicating with the remote unit  113  via downlink communication signals  116 . In the preferred embodiment of the invention, base station  101  is suitably coupled to CBSC  103 , and CBSC  103  is suitably coupled to MSC  104 , which in turn is coupled to a public switched telephone network (PSTN)  105 . 
     Referring now to FIG. 2, the base station  101  is configured as is well known to provide communications services to a plurality (typically three or six) communication sectors surrounding the base station  101 . For a first or serving sector  200 , the base station  101  has a common radio frequency (RF) front end  201  coupled to the receiver antenna array  203 . The RF front end  201  feeds baseband I, Q signals to the CDMA channel circuitry  205  and to a large storage buffer  207 . The CDMA channel circuitry  205  preferably provides for locking to at least three different received rays at least one PN chip time apart, and as such, is typical of most rake receivers utilized in CDMA systems. The CDMA channel circuitry  205  feeds the earliest PN offsets to a frequency domain searcher  209 . The channel circuitry  205  further feeds demodulated information bits to a duplicate mobile transmitter stage  211 , which re-modulates the information bits providing reference baseband I, Q signals to the searcher  209 . The searcher  209  further receives as an input the stored baseband I, Q signals from the storage buffer  207 . The searcher  209 , as will be described, provides channel parameter estimates. 
     Still referring to FIG. 2, a second or adjacent sector  202  of the base station  101  is similarly configured with an RF front end  213  coupled to an antenna array  214 , a large storage buffer  215 , a frequency domain searcher  217 , CDMA channel circuitry (not depicted) and a duplicate mobile transmitter stage (not depicted). The searcher  217  is further fed PN offset integration parameters from the searcher  209  in the serving sector  200 . It should be appreciated that reference to serving sector and adjacent sector is for the purpose of clarifying the functions performed within the base station  101 , and that the adjacent sector may act as a serving sector and vice versa without departing from the principles of the invention. 
     Turning now to FIG. 3, base station  102  is configured in a manner similar to the base station  101  serving a plurality of sectors. A single sector  300  is depicted and includes a RF front end  301  coupled to an antenna array  302 , a large storage buffer  303 , a duplicate mobile transmitter stage  305 , a frequency domain searcher  307  and CDMA channel circuitry (not depicted). Via suitable network span, the information bits from the CDMA channel circuitry  205  are fed to the duplicate mobile transmitter stage  305  along with time reference information. The duplicate mobile transmitter stage  305  re-modulates the information bits to provide reference baseband I, Q signals to the searcher  307 . The buffer  303  feeds stored baseband I, Q signals received from the remote unit  113  and for the same epoch of time corresponding to the reference baseband I, Q signals generated from the information bits. 
     One application of the present invention is for location determination of the remote unit  113  operating within the communication system  100 . It will be appreciated that the remote unit  113  may also be adapted to use the principles of the invention to determine its own location within the communication system  100 . To estimate the location of the remote unit  113 , estimates of channel parameters, such as delay, amplitude and phase must be obtained for multiple signals. As will be described, the searchers  209 ,  213  and  307  are adapted to provide such estimates. 
     With reference once again to FIGS. 2 and 3, discrete time sampled baseband I, Q representations are stored within the buffers  207 ,  215  and  307 . The samples are collected in such a way that the precise instant that they were captured is known unambiguously in terms of the system time reference. Each buffer  207 ,  215  and  307  is large, storing as much as a half-second or more of the received waveform, and may store between about 0.1 to about 2.0 seconds of the received waveform. However, one will appreciate that the invention has application where the signal sample is of substantially shorter or substantially longer duration than the aforementioned range. Data collection at the serving sector  200 , adjacent sector  202  and remote sector  300  is synchronized so that the same time-span is represented in all of the buffers  207 ,  215  and  307  at any instant. 
     While the samples are captured, they are also forwarded to the CDMA channel circuitry  205  at the serving sector  200 . The CDMA channel circuitry  205  demodulates the baseband I, Q signals of interest and produces the information bits and corresponding time information. The information bits are collected and stored in memory within the base station  101 . The channel circuitry  205  may also provide its own searcher results that may be utilized to provide a search seed and/or to narrow the search window as described below. Techniques for narrowing the search window are described in the commonly assigned U.S. patent application entitled “METHOD AND APPARATUS FOR LOCATING A REMOTE UNIT WITHIN A COMMUNICATION SYSTEM”, filed of even date herewith application Ser. No. 09/409,552 filed Sep. 30, 1999 the disclosure of which is hereby expressly incorporated herein by reference. Moreover, if a reverse-link pilot signal is available, then the described demodulation stage is not necessary. 
     Departing briefly from the foregoing discussion, as mentioned, the invention also has application to the case of a remote unit determining its location (i.e., location based on forward link signals). In such an application, pilot signals may be assumed to be available for the forward link signals of each of the base stations. Therefore, the demodulation would not be necessary. This configuration would only require the remote unit perform the downconversion, sampling and collection of complex baseband samples in a buffer for subsequent analysis, and remodulation of the known pilot sequence. 
     Returning now to the discussion in connection with FIGS. 2 and 3, when the buffers  207 ,  215  and  307  are filled and the corresponding information bits have been completely demodulated (assuming no reverse link pilot), the bits are re-modulated using the duplicate mobile transmitter stage  211  at the serving sector  200  and the duplicate mobile transmitter stage  305  at the remote sector  300  to produce reference baseband I, Q signals. The searchers  209 ,  213  and  307  process the stored waveforms and the reference waveforms to produce channel parameter estimates, such as signal delay, gain and possibly the complex channel impulse response. 
     In the preferred embodiment of the invention, the operation of each of the searchers  209 ,  213  and  307  is identical and the following description with respect to the searcher  209  is applicable to each. It will be appreciated that the functionality of the searcher  209  may be implemented as software on magnetic or optical media, firmware, a programmable gate array or an application specific integrated circuit without departing from the fair scope of the invention. With reference now to FIG. 4, within the searcher  209  multipliers  401 - 405  point-by-point multiply the baseband I, Q signal stored within the buffer  207  by the reference baseband I, Q signal over a range of delay offsets  407 - 413  from D 0  to D(S-1), where S is an initial search window length. Each row illustrated in FIG. 4 is representative of one particular delay offset evaluated by the searcher. Generally herein the term sub-group is used to refer to delay offsets; however, a sub-group may be any propagation path delay component, multipath delay component or other sub-portions of the received signal, including overlapping sub-portions, without departing from the invention. 
     The search window length, i.e., the number of sub-groups or delay offsets which require evaluation, may be shortened using searcher results from the channel circuitry  205  and/or using the techniques described in the aforementioned U.S. patent application “METHOD AND APPARATUS FOR LOCATING A REMOTE UNIT WITHIN A COMMUNICATION SYSTEM.” The delay shifts may be simply implemented by offsetting a read pointer in the buffer  207 . If the candidate PN offset Dj corresponds to an actual signal component, the resulting signal S 1   Dj(n)  should correspond to a highly over sampled low pass waveform with significant additive white noise. A hypothetical one-sided spectrum for S 1   Dj(n)  is depicted in FIG.  5 . 
     Decimators  415 - 419  decimate the product signals S 1   Dj(n)  by a factor N, and associated low pass anti-aliasing filters  421 - 425  introduce coherent averaging. The decimation may be completed efficiently (though suboptimally) by simply adding N samples together then skipping N samples. Other efficient decimation schemes are also possible. The sample rate should be decimated so that the component of the signal with the highest carrier frequency offset (including Doppler shift) is represented without frequency aliasing. For example, the maximum allowable carrier frequency offset due to non-ideal characteristics of a remote unit in accordance with the IS-95 protocol is about 300 MHZ. If the carrier is at 1900 MHZ, and the maximum velocity of the remote unit is 155 kilometers per hour (kph), the Doppler shift could be up to 269 Hz. This suggests the Nyquist frequency for decimated sample rate should be above 600 Hz. A hypothetical one-sided spectrum for S 2   Dj(n)  is depicted in FIG.  6 . 
     After every block of M decimated samples are generated, the resulting waveform of total accumulated length iM, where i is an index integer, is zero-padded (FIG. 7) to a length 2 L  by zero-padding operators  427 - 431 . A 2 L  point Fast Fourier Transform (FFT) is computed over the zero-padded signal waveform for each offset D j  and each length iM using FFT operators  433 - 437  to generate an FFT output spectrum. Because the signal is highly decimated, the FFT operators  433 - 437  should require only moderate amounts of computation and storage. The FFT output S i,Dj  for each nonzero input of length iM corresponds to iM-length coherent averaging. Each one of the 2 L  FFT outputs corresponds to coherent averaging with frequency compensation for one of 2 L  different possible carrier frequency offsets. 
     For each FFT output vector S i,Dj,  a signal to noise energy metric E i,Dj /N i  is computed by operators  439 - 443 . One of two computations may be used by the operators  439 - 443  as described below. 
     In a first preferred computation, a maximum and minimum frequency (F high  and F low , respectively) spanning the range of signal spectral components is selected by applying a threshold to the spectrum. Square energy is integrated over this whole span to produce a signal energy metric E i,Dj . The concept is illustrated in FIG.  8 . The signal component frequency range is determined by finding the maximum and minimum frequency components with energies that clear a threshold, T i . T i  is computed by multiplying the estimated noise energy N i  by a scaling constant α. The computation of N i  is described below. Signal components are only sought over a range below F max  and above F min , which represents the realistic maximum carrier frequency offsets in positive and negative directions, and thus represents a region of the FFT output spectrum that might contain signal components. For an IS-95 reverse link signal, F max  and F min  would be set at +/−600 Hz. 
     In a second preferred computation, square energy of the peak spectral components is calculated to form an energy measurement. This concept is illustrated in FIG.  9 . The peak is only sought for frequencies below F max  and above F min . 
     In each of the preferred computations, the value N i , proportional to the accumulated noise energy is computed for each signal length iM, but only at one delay offset. This offset is arbitrarily chosen to be the one with the smallest delay. The single estimate N i  is used in the ratios E i,Dj /N i  for all delay offsets. N i  is computed by summing the square energy of all frequency components above a maximum possible signal frequency, F max  and below a minimum negative frequency offset F min , in a region of the FFT output spectrum that does not contain signal components. FIGS. 7 and 8, show F min  as it would be located in an FFT output, with negative frequencies covering a range above Fs/2. 
     The energy metrics E i,Dj  are averaged to produce E i,avg  for each index i by operators  445 - 449 . The maximum energy E i,Dmax  is determined, and when for some i it exceeds the average energy for all offsets by a factor β, the search window is narrowed to a width 2W+1 (FIG.  10 ). Operator  451  accomplishes this by excluding all delay offsets D j  from further examination that fall outside the range (Dmax−W,Dmax+W). The value W should be chosen to significantly reduce the total computation required to complete the analysis, but should be wide enough to span the whole channel impulse response. The values E i,Dj  and N i  are preferably retained in memory for all surviving offsets for each index i. The despread decimated waveform for all surviving offsets are preferably retained and accumulated until the entire contents of the storage buffer are processed. 
     After an array of energy to noise ratio metrics (E i,Dj /N i ) for surviving offset candidates have been computed by increasing the nonzero lengths iM up to the full length of the storage buffer and using these as inputs to the FFT operators  433 - 437 , the best metric is determined for each offset by maximizing over i. This maximization step is illustrated in FIG.  11 . The best metric may then be mapped back to an optimal coherent averaging length, carrier frequency offset and delay offset. Providing at least one metric exceeds a threshold T 2 , the operator  453  determines the earliest offset with maximum E i /N i  exceeding the threshold T 2 . This offset is considered the earliest arriving component. 
     Now, the earliest delay offset with signal energy is known and the integration time iM that maximizes its energy-to-noise ratio is known. This offset immediately yields an estimate for time-of-arrival, which may be the only desired parameter. However, additional parameters may now be extracted from the data. The smallest delay offset satisfying the channel parameter threshold criteria is generally referred to as the “prompt ray,” i.e., the earliest arriving ray. The propagation delay of the “prompt ray” is typically the closest estimate to the distance between the mobile and the base station. 
     From the foregoing, the FFT peak value corresponding to the earliest delay yields an estimate of the complex channel gain for that signal component. The power spectrum and delay profile should be nearly identical on other antennas at the same site. Thus, using the same frequency component, delay offset and integration time iM, corresponding complex gains for the earliest component can be obtained from either adjacent sector antennas or additional phase array antennas. These gain values can be used to compute a direction-of-arrival estimate. A suitable method is described in commonly assigned U.S. patent application “METHOD AND APPARATUS FOR DETERMINING AN ANGLE OF ARRIVAL OF A TRANSMITTED SIGNAL IN A COMMUNICATION SYSTEM” by Golovin, et al., filed of even date herewith, the disclosure of which is hereby expressly incorporated herein by reference. As FIG. 2 indicates, a searcher assigned to the serving sector communicates these variables to searchers assigned to adjacent sector antennas. 
     It is possible that a channel impulse response is also desired. One motivation would be to apply interpolation and/or deconvolution in order to obtain a higher resolution estimate of mobile distance. To obtain a channel impulse response, the FFT values must be collected for a specific frequency over a range of delays and a chosen length iM. That is, a series of FFT&#39;s must be recomputed using the length that was determined to be optimal, and the outputs collected for the desired frequency. Because the despread waveforms have been retained for a search window spanning the impulse response, this should only require a small amount of additional computation. 
     Because a single frequency component is used for the described estimates, for a moving remote unit this will help isolate contributions from a single reflected path. This is because every reflection, depending on its position relative to the remote unit, will tend to produce distinct Doppler shift. The advantage is an improvement in performance when the complex gain values are used in direction-of-arrival algorithms. Moreover, the E i /N i  metrics, which estimate signal-to-noise ratio, may be directly used to measure confidence in a resulting location estimation. 
     The description of the invention, the specific details, and the drawings mentioned above, are not meant to limit the scope of the invention. For example, while the invention has been described in terms of estimating time-of-arrival for use in locating a remote unit, it may also be use to form other channel parameter estimates, such as, without limitation, signal-to-noise ratio and direction-of-arrival. It is the intent of the inventors that various modifications can be made to the invention without varying from the spirit and scope thereof, and it is intended that all such modifications come within the scope of the following claims and their equivalents.