Abstract:
A multi-output dual polarity inductive boost converter includes an inductor, a first output node, a second output node, and a switching network, the switching network configured to provide the following modes of circuit operation: a first mode where the positive electrode of the inductor is connected to an input voltage and the negative electrode of the inductor is connected to ground; 2) a second mode the negative electrode of the inductor is connected to ground and the positive electrode of the inductor is connected in sequence to one or more of the fourth and fifth output nodes; and 3) a third mode where the positive electrode of the inductor is connected to the input voltage and the negative electrode of the inductor is connected in sequence to one or more of the first, second and third output nodes.

Description:
BACKGROUND OF THE INVENTION 
     Voltage regulation is commonly required to prevent variation in the supply voltage powering various microelectronic components such as digital ICs, semiconductor memory, display modules, hard disk drives, RF circuitry, microprocessors, digital signal processors and analog ICs, especially in battery powered application likes cell phones, notebook computers and consumer products. 
     Since the battery or DC input voltage of a product often must be stepped-up to a higher DC voltage, or stepped-down to a lower DC voltage, such regulators are referred to as DC-to-DC converters. Step-down converters are used whenever a battery&#39;s voltage is greater than the desired load voltage. Step-down converters may comprise inductive switching regulators, capacitive charge pumps, and linear regulators. Conversely, step-up converters, commonly referred to boost converters, are needed whenever a battery&#39;s voltage is lower than the voltage needed to power its load. Step-up converters may comprise inductive switching regulators or capacitive charge pumps. 
     Of the aforementioned voltage regulators, the inductive switching converter can achieve superior performance over the widest range of currents, input voltages and output voltages. The fundamental principal of a DC/DC inductive switching converter is based on the simple premise that the current in an inductor (coil or transformer) cannot be changed instantly and that an inductor will produce an opposing voltage to resist any change in its current. 
     The basic principle of an inductor-based DC/DC switching converter is to switch or “chop” a DC supply into pulses or bursts, and to filter those bursts using a low-pass filter comprising and inductor and capacitor to produce a well behaved time varying voltage, i.e. to change DC into AC. By using one or more transistors switching at a high frequency to repeatedly magnetize and de-magnetize an inductor, the inductor can be used to step-up or step-down the converter&#39;s input, producing an output voltage different from its input. After changing the AC voltage up or down using magnetics, the output is then rectified back into DC, and filtered to remove any ripple. 
     The transistors are typically implemented using MOSFETs with a low on-state resistance, commonly referred to as “power MOSFETs”. Using feedback from the converter&#39;s output voltage to control the switching conditions, a constant well-regulated output voltage can be maintained despite rapid changes in the converter&#39;s input voltage or its output current. 
     To remove any AC noise or ripple generated by switching action of the transistors, an output capacitor is placed across the output of the switching regulator circuit. Together the inductor and the output capacitor form a “low-pass” filter able to remove the majority of the transistors&#39; switching noise from reaching the load. The switching frequency, typically 1 MHz or more, must be “high” relative to the resonant frequency of the filter&#39;s “LC” tank. Averaged across multiple switching cycles, the switched inductor behaves like a programmable current source with a slow-changing average current. 
     Since the average inductor current is controlled by transistors that are either biased as “on” or “off” switches, then power dissipation in the transistors is theoretically small and high converter efficiencies, in the eighty to ninety percent range, can be realized. Specifically when a power MOSFET is biased as an on-state switch using a “high” gate bias, it exhibits a linear I-V drain characteristic with a low R DS (on) resistance typically 200 milliohms or less. At 0.5 A for example, such a device will exhibit a maximum voltage drop I D ·R DS (on) of only 100 mV despite its high drain current. Its power dissipation during its on-state conduction time is I D   2 ·R DS (on). In the example given the power dissipation during the transistor&#39;s conduction is (0.5 A) 2 ·(0.2Ω)=50 mW. 
     In its off state, a power MOSFET has its gate biased to its source, i.e. so that V GS =0. Even with an applied drain voltage V DS  equal to a converter&#39;s battery input voltage V batt , a power MOSFET&#39;s drain current I DSS  is very small, typically well below one microampere and more generally nanoamperes. The current I DSS  primarily comprises junction leakage. 
     So a power MOSFET used as a switch in a DC/DC converter is efficient since in its off condition it exhibits low currents at high voltages, and in its on state it exhibits high currents at a low voltage drop. Excepting switching transients, the I D ·V DS  product in the power MOSFET remains small, and power dissipation in the switch remains low. 
     Power MOSFETs are not only used to convert AC into DC by chopping the input supply, but may also be used to replace the rectifier diodes needed to rectify the synthesized AC back into DC. Operation of a MOSFET as a rectifier often is accomplished by placing the MOSFET in parallel with a Schottky diode and turning on the MOSFET whenever the diode conducts, i.e. synchronous to the diode&#39;s conduction. In such an application, the MOSFET is therefore referred to as a synchronous rectifier. 
     Since the synchronous rectifier MOSFET can be sized to have a low on-resistance and a lower voltage drop than the Schottky, conduction current is diverted from the diode to the MOSFET channel and overall power dissipation in the “rectifier” is reduced. Most power MOSFETs includes a parasitic source-to-drain diode. In a switching regulator, the orientation of this intrinsic P-N diode must be the same polarity as the Schottky diode, i.e. cathode to cathode, anode to anode. Since the parallel combination of this silicon P-N diode and the Schottky diode only carry current for brief intervals known as “break-before-make” before the synchronous rectifier MOSFET turns on, the average power dissipation in the diodes is low and the Schottky oftentimes is eliminated altogether. 
     Assuming transistor switching events are relatively fast compared to the oscillating period, the power loss during switching can in circuit analysis be considered negligible or alternatively treated as a fixed power loss. Overall, then, the power lost in a low-voltage switching regulator can be estimated by considering the conduction and gate drive losses. At multi-megahertz switching frequencies, however, the switching waveform analysis becomes more significant and must be considered by analyzing a device&#39;s drain voltage, drain current, and gate bias voltage drive versus time. 
     Based on the above principles, present day inductor-based DC/DC switching regulators are implemented using a wide range of circuits, inductors, and converter topologies. Broadly they are divided into two major types of topologies, non-isolated and isolated converters. 
     The most common isolated converters include the flyback and the forward converter, and require a transformer or coupled inductor. At higher power, full bridge converters are also used. Isolated converters are able to step up or step down their input voltage by adjusting the primary to secondary winding ratio of the transformer. Transformers with multiple windings can produce multiple outputs simultaneously, including voltages both higher and lower than the input. The disadvantage of transformers is they are large compared to single-winding inductors and suffer from unwanted stray inductances. 
     Non-isolated power supplies include the step-down Buck converter, the step-up boost converter, and the Buck-boost converter. Buck and boost converters are especially efficient and compact in size, especially operating in the megahertz frequency range where inductors 2.2 μH or less may be used. Such topologies produce a single regulated output voltage per coil, and require a dedicated control loop and separate PWM controller for each output to constantly adjust switch on-times to regulate voltage. 
     In portable and battery powered applications, synchronous rectification is commonly employed to improve efficiency. A step-down Buck converter employing synchronous rectification is known as a synchronous Buck regulator. A step-up boost converter employing synchronous rectification is known as a synchronous boost converter. 
     Synchronous Boost Converter Operation: As illustrated in  FIG. 1 , prior art synchronous boost converter  1  includes a low-side power MOSFET switch  11 , battery connected inductor  5 , an output capacitor  8 , and “floating” synchronous rectifier MOSFET  7  with parallel rectifier diode  6 . The gates of the MOSFETs driven by break-before-make circuitry  3  and controlled by PWM controller  2  in response to voltage feedback V FB  from the converter&#39;s output present across filter capacitor  8 . Break-before-make, i.e. BBM, operation is needed to prevent shorting out output capacitor  8 . 
     The synchronous rectifier MOSFET  7 , which may be N-channel or P-channel, is considered floating in the sense that its source and drain terminals are not permanently connected to any supply rail, i.e. neither to ground or V batt . Diode  6  is a P-N diode intrinsic to synchronous rectifier MOSFET  7 , regardless whether synchronous rectifier is a P-channel or an N-channel device. A Schottky diode may be included in parallel with MOSFET  7  but with series inductance may not operate fast enough to divert current from forward biasing intrinsic diode  6 . Diode  9  comprises a P-N junction diode intrinsic to N-channel low-side MOSFET  4  and remains reverse biased under normal boost converter operation. Since diode  7  does not conduct under normal boost operation, it is shown as dotted lines. 
     If we define the converter&#39;s duty factor D as the time that energy flows from the battery or power source into the DC/DC converter, i.e. during the time that low-side MOSFET switch  4  is on and inductor  3  is being magnetized, then the output to input voltage ratio of a boost converter is proportionate to the inverse of 1 minus its duty factor, i.e. 
     
       
         
           
             
               
                 V 
                 out 
               
               
                 V 
                 in 
               
             
             = 
             
               
                 1 
                 
                   1 
                   - 
                   D 
                 
               
               ≡ 
               
                 1 
                 
                   1 
                   - 
                   
                     
                       t 
                       sw 
                     
                     / 
                     T 
                   
                 
               
             
           
         
       
     
     While this equation describes a wide range of conversion ratios, the boost converter cannot smoothly approach a unity transfer characteristic without requiring extremely fast devices and circuit response times. For high duty factors and conversion ratios, the inductor conducts large spikes of current and degrades efficiency. Considering these factors, boost converter duty factors are practically limited to the range of 5% to 75%. 
     The Need for Dual Polarity Regulated Voltages: Today&#39;s electronic devices require a large number of regulated voltages to operate, some of which may be negative with respect to ground. Some smart phones may use more than twenty-five separate regulated supplies in a single handheld, including negative bias supply needed for some organic light emitting diodes (OLEDs), displays, for biasing LCD&#39;s, and for a variety of other applications. Space limitations preclude the use of so many switching regulators each with separate inductors. 
     Unfortunately, multiple output non-isolated converters capable of generating both positive and negative supply voltage require multiple winding or tapped inductors. While smaller than isolated converters and transformers, tapped inductors are also substantially larger and taller in height than single winding inductors, and suffer from increased parasitic effects and radiated noise. As a result multiple winding inductors are typically not employed in any space sensitive or portable device such as handsets and portable consumer electronics. 
     As a compromise, today&#39;s portable devices employ only a few switching regulators in combination with a number of linear regulators to produce the requisite number of independent supply voltages. While the efficiency of the low-drop-out linear regulators, or LDOs, is often worse than the switching regulators, they are much smaller and lower in cost since no coil is required. As a result efficiency and battery life is sacrificed for lower cost and smaller size. Negative supply voltages require a dedicated switching regulator that cannot be shared with positive voltage regulators. More than one negative regulated supply voltage may be required. 
     What is needed is a switching regulator implementation capable of producing both multiple positive and negative outputs, i.e. multiple dual polarity outputs, from a single winding inductor, minimizing both cost and size. 
     SUMMARY OF THE INVENTION 
     This disclosure describes an inventive boost converter able to produce multiple independently-regulated outputs of opposite polarity, i.e. one or more positive above-ground output and one or more negative below-ground output from one single-winding inductor. A representative implementation of the dual polarity inductive boost converter includes an inductor and a switching network, the switching network configured to provide the following modes of circuit operation: 1) a first mode where the positive electrode of the inductor is connected to an input voltage and the negative electrode of the inductor is connected to ground; 2) a second mode where the negative electrode of the inductor in sequence to one or more of a first, second and third output nodes and the positive electrode of the inductor is connected in sequence to one or more of a fourth and fifth output nodes; and 3) a third mode where the positive electrode of the inductor is connected to the input voltage and the negative electrode of the inductor is connected to the either the third, fourth, or fifth output node output node. For clarification, it should be noted that said positive electrode so named because it has a higher positive potential during charging in the first mode of operation actually exhibits a negative potential during the second mode of operation. Said negative terminal of the inductor, while having a potential during magnetizing more negative than the positive terminal of the inductor, during the second and third modes of operation exhibits a more positive voltage than the inductor&#39;s other terminal. 
     The first mode of operation charges, i.e. magnetizes, the inductor to a voltage roughly equal to the input voltage. The second mode of operation transfers charge to the first or second output nodes while simultaneously transferring charge to the third, fourth, or fifth output nodes. 
     During the second mode of operation, in one embodiment of the invention after charge is transferred to the first output node, charge transfer from the inductor is diverted to the second output node while the circuit remains in its second mode of operation. During charge transfer of the second operating mode, first and second output nodes become biased to negative voltages i.e. below ground, potentials. In tandem to the sequential charging of the first and second output nodes, charge is also transferred to a third output node, followed sequentially by the charging of a fourth and optionally by a fifth output node. During charge transfer of the second operating mode, third, fourth and optionally fifth output nodes become biased to positive boosted voltages i.e. above the converter&#39;s input voltage. 
     Once the second or the fifth output node reaches its target voltage, the converter&#39;s second operating mode ends. Assuming the second output node reaches its target voltage the third mode of operation continues charging the third, fourth and fifth output nodes in sequence until the fifth reaches its target voltage. In this way, the boost converter provides five regulated outputs from a single inductor, the charging time of each output node used to determine the value of the output. 
     It will be obvious to anyone skilled in the art that this technique can be scaled to a fewer or greater number of positive and negative output channels. 
     For a second embodiment, the same basic components are used. In this case, however, the switching network provides the following modes of operation: 1) a first mode where the positive electrode of the inductor is connected to an input voltage and the negative electrode of the inductor is connected to ground; 2) a second mode where the positive electrode of the inductor is connected to the input voltage and the negative electrode of the inductor is connected to either a third, fourth or fifth output node; and 3) a third mode where the positive electrode of the inductor is connected to the first or second output node and the negative electrode of the inductor is connected to ground. 
     The first mode of operation charges the inductor to a voltage equal to the input voltage. The second mode of operation transfers charge sequentially to the third, fourth and fifth output nodes and ends when the fifth output node reaches a target voltage. The third mode of operation transfers charge sequentially to the first and second output nodes; and ends when second output node reaches its target voltage. In this way, the boost converter provides five regulated outputs from a single inductor. 
     In a third embodiment, the converter alternates between operation in accordance with the first and second embodiments depending on which output voltage requires a longer duration to reach or maintain a targeted output voltage. 
     In the first three embodiments, both positive and negative outputs are charged in tandem until one of the two outputs reaches the third state. In an alternative implementation, the two outputs are charged not simultaneously but in alternating sequence. 
     In an alternative embodiment the sequence and on time of charging the positive and negative outputs varies algorithmically in response to the output voltages staying within a predetermined tolerance range of the targeted output voltages. 
     In another embodiment the sequencing of the power MOSFETs is controlled algorithmically by a microprocessor or other programmable logic in response to feedback from one or more analog-to-digital converters monitoring the output voltages. 
     In one embodiment the power MOSFETs connecting the inductor to the positive outputs utilize either P-channel or N-channel MOSFETs with circuitry to prevent forward biasing of any diode between the MOSFETs source and drain terminals. In a preferred embodiment the MOSFET connected to the highest positive output voltage includes a source-body short and an intrinsic diode parallel to its source and drain terminals. 
     In another embodiment, the power MOSFETs connecting the inductor to the positive outputs utilize P-channel MOSFETs with the body connection tied to a fixed positive potential, preferably the most positive output voltage. 
     In one embodiment the power MOSFETs connecting the inductor to the negative outputs utilize either P-channel or N-channel MOSFETs with circuitry to prevent forward biasing of any diode between the MOSFETs source and drain terminals. In a preferred embodiment the MOSFET connected to the most negative output voltage includes a source-body short and an intrinsic diode parallel to its source and drain terminals. 
     In another embodiment, the power MOSFETs connecting the inductor to the negative outputs utilize isolated N-channel MOSFETs with the body connection tied to a fixed positive potential, preferably the most negative output voltage and with an isolation region connected to the most positive output voltage or alternatively, the converter&#39;s input 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a schematic of a prior art single output synchronous boost converter. 
         FIG. 2  is a schematic of a dual-polarity five-output synchronous boost converter as provided by the present invention. 
         FIGS. 3A-3D  show the boost converter of  FIG. 2  performing an operational sequence that implements a mode referred to as synchronous transfer. Synchronous transfer mode includes the following example successive operational phases: the inductor is magnetized ( 3 A), charge is synchronously transferred to both +V OUT1  and to −V OUT4  ( 3 B), charge is synchronously transferred to both +V OUT2  and to −V OUT5  ( 3 C), and finally charge continues to be transferred exclusively to +V OUT3  ( 3 D). 
         FIG. 4  comprises current and voltage plot of switching-waveforms characteristic of the boost converter of  FIG. 2  operating in synchronous transfer mode. 
         FIG. 5  is a flowchart for the boost converter of  FIG. 2  using synchronous transfer mode. 
         FIG. 6  illustrates the switching waveform of the boost converter of  FIG. 2  showing time multiplexed charging on the positive output nodes and their related equivalent circuits. 
         FIGS. 7  illustrates a another embodiment of the switching waveform of the boost converter of  FIG. 2  showing an alternative time multiplexed sequence charging on the positive output nodes. 
         FIG. 8  illustrates one implementation of the boost converter of  FIG. 2  using P-channel MOSFETs as synchronous rectifiers with adaptive body bias circuitry to supply the positive output nodes. 
         FIG. 9  illustrates an alternate implementation of the boost converter of  FIG. 2  using P-channel MOSFETs as synchronous rectifiers to supply the positive output nodes without requiring adaptive body bias circuitry. 
         FIG. 10  illustrates an integrated circuit cross section showing one embodiment for monolithic integrating P-channel synchronous rectifiers. 
         FIG. 11  illustrates one implementation of the boost converter of  FIG. 2  using N-channel MOSFETs as synchronous rectifiers with adaptive body bias circuitry to supply the negative output nodes. 
         FIG. 12  illustrates an alternate implementation of the boost converter of  FIG. 2  using N-channel MOSFETs as synchronous rectifiers to supply the negative outputs nodes without requiring adaptive body bias circuitry. 
         FIG. 13  illustrates an integrated circuit cross section showing one embodiment for monolithic integrating N-channel synchronous rectifiers. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     As described previously, conventional non-isolated switching regulators require one single-winding inductor and corresponding dedicated PWM controller for each regulated output voltage and polarity. In contrast, this disclosure describes an inventive boost converter able to produce multiple independently-regulated outputs of opposite polarity, i.e. one or more positive above-ground outputs and one or more negative below-ground output from one single-winding inductor. 
     Shown in  FIG. 2 , a five-output dual polarity inductive boost converter  10  comprises low-side N-channel MOSFET  11 , inductor  12 , high-side P-channel MOSFET  13 , floating positive-output synchronous rectifier  23  with intrinsic source-to-drain diode  26 , floating positive-output synchronous rectifiers  22  and  21  with no parallel source-drain diodes, floating negative-output synchronous rectifier  25  with intrinsic source-to-drain diode  27 , floating negative-output synchronous rectifier  24  with no parallel source-drain diode, output filter capacitors  31 ,  32 ,  33 ,  34  and  35  filtering outputs +V OUT1 , +V OUT2 , +V OUT3 , −V OUT4 , and −V OUT5 . Regulator operation is controlled by PWM-controller  16  including break-before-make gate buffers  17  and  18 , which control the on-time of MOSFETs  11 ,  13 ,  21 ,  22 ,  23 ,  24  and  25 . PWM controller  16  may operate at fixed or variable frequency. 
     Closed-loop regulation is achieved through independent feedback from the outputs +V OUT1 , +V OUT2 , +V OUT3 , −V OUT4 , and −V OUT5  using corresponding feedback signals V FB1 , V FB2 , V FB3 , V FB4 , and V FB5 . The feedback voltages may be scaled by resistor dividers (not shown) or other level shift circuitry as needed. Low-side MOSFET  11  includes intrinsic P-N diode  15  shown by dotted lines, which under normal operation remains reverse biased and non-conducting. Similarly, high-side MOSFET  13  includes intrinsic P-N diode  14  shown by dotted lines, which under normal operation remains reverse biased and non-conducting. High-side MOSFET  13  may be implemented using either P-channel or N-channel MOSFETs with appropriate adjustments in gate drive circuitry. 
     Unlike in conventional boost converters, in dual-polarity boost converter  10  magnetizing the inductor requires turning on both a high-side MOSFET  13  and a low-side MOSFET  11 . Inductor  12  is therefore not hard-wired to either V batt  or to ground. As a result the inductor&#39;s terminal voltages at nodes V x  and V y  are not permanently fixed or limited to any given voltage potential except by forward biasing of intrinsic P-N diodes  14  and  15  and by the avalanche breakdown voltages of the devices employed. 
     Specifically, node V y  cannot exceed one forward-biased diode drop V f  above the battery input V batt  without forward biasing P-N diode  14  and being clamped to a voltage (V batt +V f ). In the disclosed converter  10 , inductor  12  cannot drive the V y  node voltage above V batt , so that only switching noise can cause diode  14  to become forward biased. 
     Within the specified operating voltage range of the related devices, however, V y  can operate at voltages less positive than V batt  and can even operate at voltages below ground, i.e. V y  can operate at negative potentials. 
     The most negative V y  potential is limited by the BV DSS7  breakdown of high-side MOSFET  13 , a voltage corresponding to the reverse bias avalanche of intrinsic P-N diode  14 . To avoid breakdown, the MOSFET&#39;s breakdown must exceed the maximum difference between V y , which may be negative, and V batt , i.e. BV DSS7 &gt;(V batt −V y ). The maximum operating voltage range of V y  is then bounded by the breakdown and forward biasing of diode  14  given by the relation
 
( V   batt   +V   f )&gt; V   y &gt;( V   batt   −BV   DSS7 )
 
     Similarly, node V x  cannot be biased beyond one forward-biased diode drop V f  below ground without forward biasing P-N diode  15  and being clamped to a voltage V x =−V f . In the disclosed converter  10 , however, inductor  12  cannot drive the V x  node voltage below ground, so that only switching noise can cause diode  15  to become forward biased. 
     Within the specified operating voltage range of the related devices, however, V x  can operate at voltages above ground and typically operates at voltages more positive than V batt . The most positive V x  potential is limited by the BV DSS6  breakdown of low-side MOSFET  11 , a voltage corresponding to the reverse bias avalanche of intrinsic P-N diode  15 . To avoid breakdown, the MOSFET&#39;s BV DSS7  breakdown must the maximum of positive voltage of V x , which should exceed V batt , i.e. BV DSS6 &gt;V x . The maximum operating voltage range of V x  is then bounded by the breakdown and forward biasing of diode  15  given by the relation
 
 BV   DSS6   &gt;V   x &gt;(− V   f )
 
     With the V y  terminal of inductor  12  being able to operate at voltage below ground and the V x  terminal of inductor  12  being able to operate above V batt , the circuit topology of disclosed dual-polarity boost converter  10  is significantly different than conventional boost converter  1  which can only operate above ground and has its inductor hard wired to its positive input voltage. Since inductor  12  is not hard-wired to any supply rail, the disclosed dual-polarity boost converter can therefore be considered a “floating inductor” switching converter. A conventional boost converter is not a floating inductor topology. 
     Operation of the disclosed dual-polarity boost converter involves alternating between magnetizing the inductor and then transferring energy to the outputs, before magnetizing the inductor again. Energy from the inductor may be transferred to both positive and negative outputs simultaneously or in alternating fashion. 
     Whenever energy is transferred to the positive outputs, the inductor current is multiplexed sequentially to the positive outputs +V OUT1 , +V OUT2 , and +V OUT3  charging each output completely or partially to their target voltages. The sequencing of the positive outputs may occur in any order independent of the charging of the converter&#39;s negative outputs. 
     Similarly, whenever energy is transferred to the converter&#39;s negative outputs, the inductor current is multiplexed sequentially to the negative Outputs −V OUT4  and −V OUT5  charging each output completely or partially to their target voltages. The sequencing of the negative outputs may occur in any order independent of the charging of the converter&#39;s positive outputs. 
     Regardless of the algorithm employed for time multiplexing the various outputs, the first step in the operation of the disclosed dual-polarity boost converter is to store energy in, or herein to “magnetize”, the inductor, a process similar to charging a capacitor except the energy is stored in a magnetic rather an electric field. 
     Inductor Magnetizing,  FIG. 3A  illustrates operation  40  of converter  10  during the magnetizing of inductor  12 . Since inductor  12  is connected to battery input V batt  through not one, but two series connected MOSFETs, then both low-side and high-side MOSFETs  11  and  13  must be turned on simultaneously to allow current I L (t) to ramp. Meanwhile synchronous rectifier MOSFETs  21  through  25  remain off and non-conducting. The current-voltage relationship for an inductor is given by the differential equation 
     
       
         
           
             
               V 
               L 
             
             = 
             
               L 
               ⁢ 
               
                 
                   ⅆ 
                   I 
                 
                 
                   ⅆ 
                   t 
                 
               
             
           
         
       
     
     which for small intervals can be approximated by the difference equation 
     
       
         
           
             
               V 
               L 
             
             ≅ 
             
               L 
               ⁢ 
               
                   
               
               ⁢ 
               
                 
                   Δ 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   I 
                 
                 
                   Δ 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   t 
                 
               
             
           
         
       
     
     Assuming minimal voltage drop across on-state MOSFETs  11  and  13 , then V L ≈V batt  and the above equation can be rearranged as 
     
       
         
           
             
               
                 Δ 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 I 
               
               
                 Δ 
                 ⁢ 
                 
                     
                 
                 ⁢ 
                 t 
               
             
             = 
             
               
                 
                   V 
                   L 
                 
                 L 
               
               ≈ 
               
                 
                   V 
                   batt 
                 
                 L 
               
             
           
         
       
     
     which describes for short magnetizing intervals the current I L (t) in inductor  12  can be approximated as a linear ramp of current with time. For example as shown in graph  90  of  FIG. 4 , during the interval between to and t 1  the current I L  ramps linearly  91  from some non-zero current at time t 0  toward a peak value  92  at time t 1 , the end of the magnetizing operating phase. The energy stored in inductor  12  at any time t is given by 
     
       
         
           
             
               
                 E 
                 L 
               
               ⁡ 
               
                 ( 
                 t 
                 ) 
               
             
             = 
             
               
                 
                   LI 
                   2 
                 
                 ⁡ 
                 
                   ( 
                   t 
                   ) 
                 
               
               2 
             
           
         
       
     
     reaching its peak E L (t 1 ) just before its current is interrupted by switching off one or both MOSFETs  11  and  13 . As shown in graphs  60 ,  75  and  90  of  FIG. 4 , during magnetizing the current I 6  in low-side MOSFET  11  and the current  17  in high-side MOSFET  13  are identical and equal to the inductor current I L  so that in the interval t 0  to t 1 ,
 
 I   6 ( t )= I   7 ( t )= I   L ( t )
 
     At current I 6 (t), a small voltage drop V DS2(on)  appears across series-connected low-side N-channel MOSFET  11 . Operating in its linear region and carrying current I L (t) with an on-state resistance of R DS6(on)  the voltage V x  is given by
 
 V   x   =V   DS6(on)   =I   L   ·R   DS6(on)  
 
     as shown by line  61  in graph  60  of  FIG. 4 . For low on-resistances, typically a few hundred milliohms or less, then V x  is approximately equal to ground potential, i.e. V x ≈0. Similarly, a small voltage drop V DS7(on)  also appears across series-connected high-side P-channel MOSFET  13 . Operating in its linear region at a current I L (t) with an on-state resistance of R DS7(on)  the voltage V y  is then given by
 
 V   y   =V   batt   −V   DS7(on)   =V   batt   −I   L   ·R   DS7(on)  
 
     as shown by line  52  in graph  75  of  FIG. 4 . For low on-resistances, then V y  is approximately equal to the battery potential, i.e. V y ≈V batt . 
     Given that V x ≈0 and V y ≈V batt  then the approximation V L =(V y −V x )≈V batt  is a valid assumption. Accordingly, the ramp in inductor current shown in graph  90  can, as described previously, therefore be approximated as a straight line segment with a slope (V batt /L). Furthermore assuming the voltage +V OUT3  across capacitor  31  is above ground and the voltage −V OUT5  across capacitor  35  is below ground, then +V OUT3 &gt;V x  and V y &gt;−V OUT5  so that P-N diodes  26  and  27  are both reverse biased and non-conducting. 
     Synchronous Energy Transfer to Dual Polarity Outputs: After magnetizing inductor  12 , in the synchronous transfer algorithm both low-side and high-side MOSFETs  11  and  13  are turned off simultaneously, as shown at time t 1  of  FIG. 4 . Interrupting the I 7  current in high-side MOSFET  13  and the I 6  current in low-side MOSFET  11  causes the inductor&#39;s V x  terminal to fly up to a positive voltage  63  greater than V OUT1 , forward biasing diode  26 . Before any appreciable energy flows to capacitor  33  and output +V OUT3 , MOSFET  21  turns on and reroutes the current to capacitor  31  and output +V OUT1  as shown in  FIG. 3B . The voltage V X  then settles to a voltage  63  only slightly above +V OUT1 , the product of the inductor current and the on-resistance of synchronous rectifier MOSFET 21 . i.e. I L R DS1 . 
     As shown in  FIG. 3B  synchronous to inductor  12  transferring energy to capacitor  31 , the same inductor also charges capacitor  34  to a negative potential. Specifically at time t=t 1  as shown in graph  75  of  FIG. 4  the inductor&#39;s V y  terminal voltage flies down to a below-ground voltage  80 , a voltage more negative than −V OUT5 , momentarily forward biasing diode  27 . Before any appreciable energy is transferred to capacitor  35 , MOSFET  24  is turned on reroutes the inductor&#39;s current to the negative voltage output −V OUT4 . The voltage V Y  then settles to a voltage  81  only slightly below −V OUT4 , the product of the inductor current and the on-resistance of synchronous rectifier MOSFET  24 . i.e. I L R DS4 . 
     During the transition at time t 1 , break-before-make circuits  17  and  18  prevents synchronous rectifier MOSFETs  21  and  24  from turning on and momentarily shorting out filter capacitors  31  and  34 . Without MOSFET conduction, diodes  26  and  27  carry the inductor current I L  and exhibit a forward-biased voltage-drop V f . In the absence of stray capacitance in the circuit, the instantaneous voltage  62  on V x  during the BBM interval is then equal to (V OUT3 +V f ). The instantaneous voltage  80  on V y  is similarly equal to (−V OUT5 −V f ). If however, the BBM interval is sufficiently short, stray capacitance filters the voltage spikes  62  and  80  on V X  and V Y  to a magnitude slightly larger than +V OUT1  and −V OUT4 . As a result, no significant energy flows to capacitors  33  and  35  during the BBM interval and the output voltages +V OUT3  and −V OUT5  are not disturbed. 
     Since no significant power flows through them, the function of diodes  26  and  27  is to act as a clamp to prevent short unwanted spikes and related noise on the V X  and V Y  and to avoid driving MOSFETs  11  and  13  into a potentially harmful unclamped-inductively-switched avalanche condition. In graphs  60  and  75  theses short BBM intervals are represented as temporary V X  voltage transients  62 ,  68 ,  69 ,  70  and temporary V Y  voltage transients  80 ,  84  and  85 . 
     After the BBM interval at time t 1 , the positive and negative outputs +V OUT1  and −V OUT4  charge simultaneously during which inductor  12  essentially decouples the voltage at nodes V x  and V y  allowing them to act independently during the time energy is transferred to the loads and to output capacitors  31  and  34  through the synchronous rectifier MOSFETs  21  and  24 . 
     In one method, the condition shown in schematic  3 B should continue until the voltage on either capacitor  31  or  34  comes into a specified tolerance range. The tolerance range of the target voltage is determined by the controller in response to the feedback signals V FB1  and V FB4 . Using analog control, the PWM controller  16  includes an error amplifier, a ramp generator, and a comparator to determine when to shut off each synchronous rectifier. Using digital control, this decision can be made by logic or software according a specified algorithm, depending on the load current demand and capacitor voltage on each output. 
     For example when the positive output +V OUT1  reaches its target voltage at time t 2 , positive synchronous rectifier MOSFET  21  is turned off discontinuing charging of capacitor  31 . After a BBM interval MOSFET  22  is turned on and capacitor  32  then commences charging of output voltage +V OUT2 . The voltage V X  then increases to (+V OUT2 +I L R DS2 ) as shown by line  65  in graph  60 . 
     At a later time, i.e. at time t 3 , the negative output −V OUT4  reaches its specified tolerance range, synchronous rectifier MOSFET  24  is turned off. After a BBM interval, MOSFET  25  is turned on and capacitor  35  commences charging of output −V OUT5 . Charging of capacitor  32  and voltage +V OUT2  continues. This condition  50  from time t 4  to t 5  is illustrated in  FIG. 3C  during which the voltage V X  then increases to (+V OUT3 +I L R DS3 ) as shown by line  75  in graph  60  and where V Y  increases to (−V OUT5 −I L R DS5 ) as shown by line  83  in graph  75 . 
     Synchronous Energy Transfer to One Polarity Output: Depending on load conditions either positive or negative polarity outputs may become fully charged within their tolerance ranges first. Once either output reaches its specified output voltage, the converter is again reconfigured to discontinue charging of the fully charged polarity but continue charging the output capacitors not yet within the tolerance range its specified voltage target. 
     For example, if at a time t 5  the negative output −V OUT5  reaches its target voltage before +V OUT3 , then the first action is to turn off synchronous rectifier MOSFET  25 , and disconnect capacitor  35  from over charging. After BBM interval  59  is completed, high-side MOSFET  13  is turned-on and V y  jumps to a voltage of V batt −I L ·R DS7(on)  shown by line  56  in graph  60 . During the hand-off at time t 5 , inductor current I L  is diverted from I 5  to I 7  in the transition shown by point  84  in graph  74 . Current I 3  however remains unchanged. 
     This condition is shown in circuit  55  of  FIG. 3D  where the current path of I L  flows from V batt  through conducting high-side MOSFET  13 , inductor  12 , and on-state positive synchronous rectifier  23  so that I L =I 7 =I 3 . Capacitor  33  therefore continues to charge even though charging of capacitor  35  has stopped. With V y  biased near V batt  and −V OUT5  below ground P-N diode  27  remains reversed biased and non-conducting. 
     The operating phase of circuit  35  is maintained until +V OUT3  reaches its target voltage at time T. Once +V OUT3  is at its target voltage, positive synchronous rectifier MOSFET  23  is turned off and for the break-before-make duration t BBM    68 , diode  26  carries the inductor current. 
     Once however the BBM interval  68  is completed low-side MOSFET  11  is turned on, current is diverted from I 3  to I 6  and inductor  12  begins a new cycle of being magnetized returning to the state shown in circuit  40 . Having completed the cycle, the total time is described as the period T which will vary depending on load current. This period is determined by the magnetizing duration and the positive or negative charge transfer phases which ever is longer. 
     The example given in  FIG. 3D  described a case where the negative Output −V OUT5  reached its target voltage before the positive Output +V OUT3 . The converter also accommodates the opposite scenario, i.e. when the positive voltage hits its point of regulation first. 
     As shown in graph  90  in the synchronous transfer method, the inductor  12  must supply all five outputs with charge over an interval from t 1  to T with the inductor current decaying  93  from a peak current  92  to a minimum value  94  before the cycle repeats. For conservation of energy, the magnetizing energy during the interval t 0  to t 1  must equal the energy delivered in the remainder of the period. 
     State Diagram of Synchronous Charge Transfer The algorithm and state diagram for synchronous transfer  100  is illustrated in  FIG. 5 . As shown the initial state  110  involves magnetizing the inductor, then simultaneously operations  101  and  106  power both negative and positive outputs by turning off the high side and low side MOSFETs  11  and  13  and time multiplexing the synchronous rectifiers. In flow  101 ,  102 ,  103  the positive polarity outputs +V OUT51 , +V OUT2 , and +V OUT3  are charged  111  sequentially as shown or in any sequence. In tandem to charging the positive outputs, in flow  106 ,  107  the converter&#39;s negative outputs are charged in any sequence. 
     The charging of the converter&#39;s positive outputs is controlled by the low-side MOSFET  11  connected between ground and V X . Turning off MOSFET  11  commences charging according to multiplexed sequence  101 ,  102 , and  103 . To terminate positive charging  104 , low side MOSFET  11  must be turned back on in state  112 . The result is conditional. If high side MOSFET  13  is already on, then turning on low side MOSFET  112  will re-initiate  105  magnetizing inductor  12  shown by state  110 . If the high side MOSFET is still off, i.e. if negative charging sequence  106 ,  107  is still ongoing, then the positive loop will wait in condition  112 . 
     Similarly, the charging of the converter&#39;s negative outputs is controlled by the high-side MOSFET  13  connected between V batt  and V Y . Turning off MOSFET  13  commences charging of the negative outputs  113  according to multiplexed sequence  106  and  107 . To terminate negative charging  108 , high side MOSFET  13  must be turned back on in state  114 . The result is conditional. If low side MOSFET  11  is already on, then turning on low side MOSFET  115  will re-initiate  109  magnetizing inductor  12  shown by state  110 . If the low side MOSFET is still off, i.e. if the positive charging sequence  101 ,  102 ,  103  is still ongoing, then the negative loop will wait in condition  114 . 
     In the synchronous transfer method, both loops in algorithm  100  occur simultaneously, the longer loop sets the duration of the repeated interval, i.e. the converter&#39;s period T. For example if the negative charge transfer sequence  106 ,  107 ,  108  occurs in a shorter time than the positive loop  101 ,  102 ,  103 ,  104 , the negative loop will wait at state  114  with its high side MOSFET on until the positive loop reaches state  112 . When state  112  finally is reached, then the converter returns to the starting condition by paths  105  and  109  simultaneously. 
     Conversely, if the positive charge transfer sequence  101 ,  102 ,  103 ,  104  occurs in a shorter time than the negative loop  106 ,  107 ,  108 , the positive loop will wait at state  112  with its low side MOSFET on until the negative loop reaches state  114 . When state  114  finally is reached, then the converter returns to the starting condition by paths  105  and  109  simultaneously. 
     In the disclosed approach, charging is synchronous because both high side and low side MOSFETs are turned off simultaneously thereby immediately forcing both sides of the inductor V X  and V Y  to exhibit voltage transients an charging their respective outputs. 
     An alternative approach is to alternate between the positive and the negative loops, first by magnetizing the inductor, turning off only the low side MOSFET and completing positive output loop  101 ,  102 ,  103 ,  104 ,  105 , returning to magnetizing state  110 , turning off only the high side MOSFET, completing negative output loop  106 ,  107 ,  108 ,  109 , and then repeating the entire process. 
     Voltage Regulation of the Dual-Polarity Multiple Output Regulator: Operation of the dual polarity boost converter requires turning on both high-side and low-side MOSFETs  13  and  11  to magnetize inductor  12  and then shutting off these MOSFETs to transfer energy to the converters outputs. In the synchronous energy transfer algorithm, both aforementioned high-side and low-side MOSFETs are shut off simultaneously starting the transfer of energy from the inductor to both outputs simultaneously. 
     Despite being charged synchronously, independent regulation of the positive and negative outputs are determined by the duration of energy transfer to each output. Specifically, by controlling the off-time of the low-side and high-side MOSFETs  11  and  13  and the relative on-time for each of the synchronous rectifier MOSFETs through feedback, the various positive and negative output voltages may be independently regulated from a single inductor. 
     Time Multiplexed Sequencing In the disclosed invention, any time multiplexed sequence may be used for producing the multiple positive or multiple negative outputs. For example in  FIG. 6 , the positive outputs are charged in successive monotonic sequence starting with the lowest output voltage +V OUT1 , progressing to the second output +V OUT2 , and finally charging the highest output voltage +V OUT3 . The graph illustrates the initial charging of the output capacitors during start up, not just steady state operation. 
     More specifically from time t 0  to t 1  sub-circuit  140  illustrates the low-side MOSFET  11  is on and the synchronous rectifiers are off. All three output voltages  131  in graph  130  are at zero and V X  in graph  120  has a potential  121  equal to the voltage drop I L R DS(on)  across the conducting low side MOSFET  11 . 
     After MOSFET  11  is shut off at time t 1 , synchronous rectifier  21  shown in sub-circuit  141  is turned on and +V OUT1  ramps  132  to its target voltage V′ OUT1 . At the same time diode  26  becomes forward biased ramping +V OUT3  to value of (+V OUT1 −V f ). The inductor node voltage V X  drives the outputs voltage up with a ramp  122  limited by the charging of the converter&#39;s output capacitances. During this period, +V OUT2  remained at ground. 
     At time t 2  MOSFET  21  is turned off and MOSFET  22  is turned on as shown in sub-circuit  142 . As V x  continues  123  to rise, the output voltage +V OUT2  charges to a target voltage V′ OUT2  and the forward biasing of diode  26  continues to ramp output +V OUT3  to value of (+V OUT2 −V f ) as shown by line  135 . With MOSFET  21  off, the output +V OUT1  remains constant at its targeted value V′ OUT1 . 
     At time t 3  MOSFET  22  is turned off and MOSFET  23  is turned on as shown in sub-circuit  143 . As V x  continues  124  to rise, the output voltage +V OUT3  charges  138  to a target voltage V′ OUT3  with MOSFET  23  shunting forward biased diode  26 . With MOSFETs  21  and  22  off, the lower voltage outputs +V OUT1  and +V OUT2  remain constant at or near their targeted values V′ OUT1  and V′ OUT2 . 
     After time T, the circuit enters steady state operation with only small changes in the converter&#39;s outputs manifest thereafter. 
     An alternative multiplexing sequence, one where the V X  voltage does not ramp monotonically, is illustrated in  FIG. 7  where graph  150  illustrates V X  and graph  160  illustrates the various output voltages. Specifically, before time t 1 , the inductor is magnetizing, V X  is biased near ground, and all three outputs are zero. 
     At time t 1 , the low side MOSFET is turned off and MOSFET  23  is turned on, whereby the converter&#39;s highest output-voltage +V OUT3  rises  162  toward its target value V′ OUT3  driving by V X  voltage  152 . Other outputs +V OUT1  and +V OUT2  remain grounded. 
     At time t 2 , the converter&#39;s lowest output voltage +V OUT1  is next powered ramping  163  in proportion to V X    153  toward a target value of V′ OUT1 . The output +V OUT2  remains grounded. Because the output +V OUT3  is fully charged, P-N diode  26  becomes reverse biased. 
     At time t 3  the converter&#39;s median output voltage +V OUT2  is next powered ramping  164  in proportion to V X    154  toward a target value of V′ OUT2 . The outputs +V OUT1  and +V OUT1  remain at their previous values. Because the output +V OUT3  is fully charged, P-N diode  26  remains reverse biased. 
     So the disclosed converter&#39;s initial multiplexing sequence can be implemented without concern for special sequencing for monotonic operation. 
     In the example of circuit  10  operation, in steady state operation only one positive and one negative synchronous rectifier are turned on at a time. Specifically as shown whenever low-side MOSFET  11  is biased off and the voltage at node V x  increases, only one positive-output connected synchronous rectifier MOSFET, either MOSFET  21 ,  22  or  23  is turned on at the same time. Similarly, whenever high-side MOSFET  13  is turned off and V Y  flies negative, only one negative-output connected synchronous rectifier MOSFET, either MOSFET  24  or  25  is turned on at the same time. Turning on more than one synchronous rectifier would in effect short out the affected outputs and cause the voltages to equilibrate. 
     For example simultaneously turning on synchronous rectifier MOSFETs  21 ,  22  and  23  will cause the voltage to equilibrate among capacitors  31 ,  32  and  33 . The adverse effect of voltage equilibration is that charge flows “backwards” from the highest output voltage capacitor into the lowest voltage capacitor lowering converter efficiency and increasing output ripple or causing noise spikes on the affected outputs. If, during such as condition V OUT3 &gt;V OUT2 , then by simultaneously turning on MOSFETs  23  and  22 , capacitor  32  would be charged by a combination of both inductor  12  current and capacitor  33  current causing V OUT2  to rise and V OUT3 . to drop in voltage. Energy redistribution among the filter capacitors is less efficient than supplying new charge to the outputs from inductor  12  current as need. 
     Similarly, simultaneously turning on synchronous rectifier MOSFETs  24  and  25  will cause the voltage to equilibrate among capacitors  34  and  35 . The adverse effect of voltage equilibration is that charge flows “backwards” from the highest, i.e. the most negative, output voltage capacitor into the lowest voltage capacitor. Such currents lower converter efficiency and increase output ripple and noise spikes on the affected outputs. If, during such as condition −V OUT5 &lt;−V OUT4 , then by simultaneously turning on MOSFETs  24  and  25 , capacitor  34  would be charged by a combination of both inductor  12  current and capacitor  35  current causing −V OUT5  to rise and −V OUT3 . to drop, i.e. be come less negative, in voltage. Energy redistribution among the filter capacitors is less efficient than supplying new charge to the outputs from inductor  12  current as need. 
     Since the positive and negative outputs are connected to opposite terminals V X  and V Y  of inductor  12 , the selection or sequence of positive-output connected synchronous rectifiers places no limitations on which negative-output connected synchronous rectifier MOSFET is conducting in tandem, or vice versa. As long as only one positive-output connected MOSFET and only one negative output connected MOSFET are conducting, no intra-capacitor charge redistribution will occur and efficiency will not be lost. 
     Although turning on multiple synchronous rectifiers of the same polarity does not necessarily damage the devices of circuit  10  or prevent regulation, it offers no technical merit and generally suffers a number of the aforementioned problems. 
     One condition, however, allows the outputs to be shorted by multiple conducting synchronous rectifiers without causing charge redistribution losses in the capacitors. This condition occurs during start up when the capacitors are being charged for the first time. As long as the capacitor voltages are similar all the synchronous rectifiers may be turned on in tandem and allow the inductor to simultaneously charge every positive and negative output. This process expedites turn on and ramp up of the converter to its steady state. Once a given output capacitor reaches its target range, it is disconnected from the inductor while the other channels continue to charge. Once disconnected the isolated capacitor will quickly exhibit its own unique output voltage and thereafter may be not be reconnected in parallel with the other outputs without charge redistribution losses occurring. 
     Other Features of the Dual Polarity Multi-Output Converter One feature of the disclosed converter  10  is that since the inductor is floating, i.e. not permanently connected to a supply rail, turning on either the high-side or low-side MOSFETs  11  and  13  but not both can force the voltage at V y  or V x  without magnetizing or increasing the current in inductor  12 . This is not possible for a conventional boost converter like the one in  FIG. 1  where a single MOSFET both controls the V x  voltage but also causes current conduction, magnetizing the inductor. In other words in a conventional converter, controlling the inductor voltage also causes additional and sometimes unwanted energy storage. In the disclosed converter, either V x  or V y  can be forced to a supply voltage without magnetizing the inductor. 
     Another consideration is the output voltage range of conventional boost converter  1 . If a P-N diode  6  is present across a synchronous rectifier MOSFET, the minimum output voltage for the boost converter&#39;s output is necessarily V batt , because the diode forward biases pulling the output up to V batt  as soon as power is applied to the regulator&#39;s input terminals. In the disclosed dual output converter, the circuit path from V X  to +V OUT1  or to +V OUT2  includes MOSFETs with no source-drain parallel P-N diodes, allowing +V OUT1  or +V OUT2  to regulate a voltage less than V batt , a feature not possible with a conventional boost converter topology. 
     So while boost converters can only step up voltage, the disclosed converter produces a positive output voltage that can be less than, equal to or greater than the battery voltage, and is therefore not restricted to operation only above V batt . Adapting a boost converter&#39;s topology for step-down voltage regulation is the subject of a related patent by Richard K. Williams entitled “High-Efficiency Up-Down and Related DC/DC Converters” (now U.S. patent application Ser. No. 11/835,809) and is included herein by reference. 
     In a related disclosure entitled “Dual-Polarity Multi-Output DC/DC Converters and Voltage Regulators” (now U.S. patent application Ser. No. 11/890,818) by Richard K. Williams, the application of a time-multiplexed-inductor in both positive and negative output boost converters is described and is incorporated herein by reference. 
     Multiplexer Implementation The disclosed dual polarity multi-output converter requires the use of MOSFETs free from parasitic source-to-drain diodes. In order to implement power MOSFETs without intrinsic source to drain diodes, a number of methods are herein disclosed. Once such method illustrated in sub-circuit  180  of  FIG. 8  comprises P-channel MOSFETs with integral body bias generator circuitry. 
     As shown P-channel MOSFETs  21  and  22  include associated body bias generator or BBG circuits  191 A and  181 A respectively 
     BBG circuit  191 A comprises cross coupled MOSFETs  192  and  193  sharing a common body connection to main P-channel MOSFET  21 , the body which represents the cathode of intrinsic diodes  194  and  195 . In its integrated version BBG circuit  191 A contained in an N-type well or tub may include a parasitic diode  191 B to ground. BBG circuit  181 A is of similar construction to  191 A. 
     Operation of BBG circuit selectively shunts diodes  194  and  195  so that neither diode can become forward biased and carry current regardless of the polarity of the source-drain terminals of MOSFET  21 . For example if V OUT1 &gt;V X , then the gate of P-channel MOSFET  193  is more negative than its other terminals so that P-channel  193  is on, shunting P-N diode  195 . The body of P-channel MOSFET  21  is therefore connected to V OUT1 , the most positive device potential. Consequently, diode  194  remains reversed biased and non-conducting. P-channel  192 , with its gate to the most positive potential also remains off. Since the device and circuit are symmetric the argument applies equally in both polarities. As a result BBG circuit  191 A makes MOSFET  21  appear as if it has no parallel P-N diode that can ever become forward biased. 
     In circuit  180 , while MOSFETs  21  and  22  use BBG circuits  191 A and  181 A to prevent body diode conduction, MOSFET  23  connected to the most positive output voltage +V OUT3 , does not require a BBG circuit. In fact parallel diode  26  is important as insurance to prevent V x  from large voltage spikes. 
     In integrated form the N-well or epitaxial layers  181 A and  191 A forming the body of the P-channel MOSFETs also form parasitic diodes  181 B and  191 B to the surrounding P-type substrate. Since V X  is always positive, these diodes remain reverse biased during normal operation of body bias generator circuitry  180 . 
     A similar approach to circuit  180  can be employed using N-channel power MOSFETs in place of P-channel devices but the gate drive circuitry must be modified accordingly for bootstrap circuitry with a floating bootstrap capacitor. Generally a special wafer fabrication process is required to isolate N-channel MOSFETs from a surrounding P-type substrate. A P-type substrate is common in most wafer fabrication processes, especially in conventional CMOS processes. 
     Alternate P-channel solution  200  shown in  FIG. 9  utilizes the knowledge that the output +V OUT3  is the most positive potential and can be used to avoid diode conduction in MOSFETs  21  and  22 . As such P-channel MOSFET  21  includes intrinsic body diodes  194  and  195  with cathode and N-type well region  191 A electrically tied to +V OUT3 , the circuit&#39;s most positive potential. Similarly P-channel MOSFET  22  includes intrinsic body diodes  184  and  185  with cathode and N-type well region  181 A electrically tied to +V OUT3 , the circuit&#39;s most positive potential. Because N-type wells  181 A and  191 A are biased at the highest potential, i.e. at +V OUT3 , then in an integrated form parasitic diodes  181 B and  191 B to the surrounding P-type substrate remain off and reversed-biased during normal circuit operation. One disadvantage of circuit  200  compared to the BBG method of circuit  180  is the reverse biased source to body bias will result in an increase in threshold voltage and on-resistance. 
     Regardless of the circuit, the P-channel devices shown in schematics  180  and  200  can be integrated using standard CMOS or preferably realized in a process that provides a heavily doped high, i.e. high concentration, buried layers. One such implementation illustrated in cross section  220  of  FIG. 10  utilizes a deep implanted N-type region  222  overlapping the bottom of N-type well  223 . In the implementation shown well  223  and deep implant  222  are formed conformal to LOCOS oxide  233 . The P-channel MOSFET includes P+ source drain regions  224 , P-type lightly doped drain extension  225 , sidewall spacer  229 , polysilicon gate  228 , gate silicide  227 , contact barrier metal  235 , 1 st  layer metal  231  and second layer metal  232 . This concept, referred to herein as a variable gate width switching converter, is described in prior art U.S. Pat. No. 5,973,367 by Richard K. Williams and in another implementation in U.S. Pat. No. 7,026,795 by John So. 
     As an example, in cross section  220  the N-type well shared by MOSFETs  21 ,  22  and  23  are biased to the positive output potential +V OUT3 . P-channel  21 , not shown in cross section  229  can also be integrated into the same shared well. If a BBG circuit such as shown in  FIG. 8  is required, however MOSFETs  21 ,  22  and  23  must each employ their own separate wells and cannot share a common one. 
     The same BBG circuit technique can be applied for N-channel MOSFETs used as synchronous rectifiers for the converter&#39;s negative outputs. As shown in circuit  250  of  FIG. 11 , MOSFET  24  includes cross coupled N-channel MOSFETs  254  and  255  to selectively shunt intrinsic diodes  253  and  252 . Since the body of N-channel  24  is P-type it must be isolated from the P-type substrate in order to monolithically integrate circuit  250 . Isolation of N-channel MOSFETs generally requires a special wafer fabrication process whereby a N-type isolation layer  257 A surrounding and enclosing the MOSFET forms a reversed biased diode  257 B with the surrounding P-type substrate and a reversed biased diode  257 C with said enclosed P-type well. N-type isolation layer  257 A is biased at voltage more positive than its operating voltages, e.g. to +V batt  or +V OUT3  to prevent forward biasing of the isolation region to its surroundings. Floating N-channel MOSFET  25  also requires isolation since it is not ground connected, even though it contains source to drain diode  27  and a source-body short. 
     An alternative approach shown in circuit  270  of  FIG. 12 , biases the P-type body of N-channel MOSFET  24  to the most negative circuit potential −V OUT5 . The floating device still requires isolation from the P-type substrate using a specialized process with isolated devices. Formed in a P-well surrounded by an N-type isolation layer  271 A, the isolation is generally biased to a potential V ISO  more positive than the operating voltage range of the device. Positive bias supplies may comprise the V batt  input or a positive output such as +V OUT3 . Properly biased, N-type isolation layer  271 A forms a reversed biased diode  271 B with its surrounding P-type material and also forms a reverse biased diode  271 C with the P-well it encloses. 
     One such implementation is shown in cross section  300  with a deep implanted N-isolation region  302  surrounding a P-type well  303  containing two N-channel MOSFETs for producing outputs −V OUT4  and −V OUT5 . The isolation is independently biased to a potential equal to or more positive that V batt . The N-channel MOSFET includes N+ source drain regions  305 , N-type lightly doped drain extension  306 , sidewall spacer, gate oxide  307 , polysilicon gate  309 , gate silicide  310 , contact barrier metal  313 , 1 st  layer metal  312  and second layer metal (not shown). The process to fabricate such a device is also described in U.S. Pat. No. 6,855,985 “Modular Bipolar-CMOS-DMOS Analog Integrated Circuit &amp; Power Transistor Technology” by Richard K. Williams et al. incorporated herein by reference.