Abstract:
Disclosed is a synchronization tracking circuit for synchronizing the phase of a despreading code sequence on a receiving side to the phase of a spreading code sequence on a transmitting side. The synchronization tracking circuit has a DLL circuit for performing synchronization tracking by DLL (Delay Locked Loop) control, and an interference-component estimation unit for estimating an interference component inflicted by another path upon a prescribed path of interest among multiple paths. The DLL circuit, which has an interference elimination unit, executes DLL control based upon a signal from which the interference component from the other path has been eliminated and causes the phase of the despreading code sequence on the receiving side to be synchronized with and track the phase of the spreading code on the transmitting side.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    This invention relates to a synchronization tracking circuit for causing the phase of a despreading code sequence on a receiving side to follow up the phase of a spreading code sequence on a transmitting side. More particularly, the invention relates to a synchronization tracking circuit, which is used in the field of CDMA mobile communications employing spread spectrum, for exercising DLL (Delay Locked Loop) control in such a manner that a despreading code sequence on the receiving side will not shift in time with respect to a receive signal for which acquisition of synchronization has succeeded.  
           [0002]    In a CDMA (Code Division Multiple Access) mobile communications system using spread spectrum, the transmitting side transmits information upon spreading the information using a spreading code sequence, and the receiving side demodulates the transmit information upon despreading the signal from the transmitting side using a despreading code sequence that is identical with the spreading code sequence.  
           [0003]    [0003]FIG. 12 is a block diagram illustrating the construction of a CDMA receiver. The receiver includes a radio unit  1  that subjects a high-frequency signal received by an antenna ANT to a frequency conversion (RF→IF conversion) to obtain baseband signals. A quadrature detector  2  subjects the baseband signals to quadrature detection and outputs in-phase component (I component) data and quadrature component (Q component) data. The quadrature detector  2  includes a receive carrier generator  2   a , a phase shifter  2   b  for shifting the phase of the receive carrier by π/2, and multipliers  2   c ,  2   d  for multiplying the baseband signals by the receive carrier and outputting the I-component and Q-component signals. Low-pass filters (LPF)  3   a ,  3   b  limit the bands of the output signals and A/D converters  4   a ,  4   b  convert the I- and Q-component signals, respectively, to digital signals. The digital signals are input to a searcher  5  and to each of fingers  6   1  to  6   4.    
           [0004]    If a direct sequence signal (DS signal) that has experienced multipath effects is input to the searcher  5 , the latter executes autocorrelation processing using a matched filter (not shown), detects multipath and inputs, to the fingers  6   1  to  6   4 , despreading-start timing data τ 0  to τ 3 , respectively, and delay-time adjustment data for the respective paths. Each of the fingers  6   1  to  6   4  has a despreading code generator  6   a  for generating a code sequence identical with the spreading code sequence on the transmitting side based upon the timing data τ 0  to τ 3  that enters from the searcher  5 . More specifically, the searcher  5  detects the phase of the transmitting-side spreading code (referred to as “synchronization capture”) at a precision of within one chip, and the despreading code generator  6   a  generates a despreading code sequence, which is for performing despreading on the receiving side, in sync with the detected phase. A DLL (Delay Locked Loop) circuit  6   b  exercises control (referred to as “synchronization tracking”) in such a manner that the despreading code sequence on the receiving side will not develop a time shift with respect to a receive signal for which synchronization has been captured even if the receive signal undergoes a change in phase owing to the effects of modulation and noise, etc.  
           [0005]    Each finger further includes a despreader/delay-time adjustment unit  6   c  for performing dump integration by subjecting a direct wave or a delayed wave that arrives via a prescribed path to despread processing using a code identical with the spreading code, and for subsequently applying delay processing that conforms to the path and outputting a pilot signal (reference signal) and information signal; a phase compensator (channel estimation unit)  6   d  for averaging voltages of the I and Q components of the pilot signal over a prescribed number of slots and outputting channel estimation signals It, Qt; and a synchronous detector  6   e  for restoring the phases of despread information signals I′, Q′ to the original phases based upon a phase difference θ between a pilot signal contained in a receive signal and an already existing pilot signal. More specifically, the channel estimation signals It, Qt are cosine and sine components of the phase difference θ, and therefore the synchronous detector  6   e  demodulates the receive information signal (I,Q) (performs synchronous detection) by applying phase rotation processing to the receive information signal (I′,Q′) in accordance with the following equation using the channel estimation signal (It,Qt):  
         (         I           Q         )     =       (         It       Qt             -   Qt         It         )          (           I   ′               Q   ′           )                             
 
           [0006]    A rake combiner 7 combines signals output from the fingers  6   1  to  6   4  and outputs the combined signals to an error correction decoder  8  as a soft-decision data sequence. The error correction decoder  8  applies error correction processing, demodulates the transmit information and outputs the demodulated signal.  
           [0007]    DLL circuit  
           [0008]    As mentioned above, a CDMA receiver has a searcher for detecting the phase of the transmitting-side spreading code (referred to as “synchronization capture”) at a precision of within one chip, after which a despreading code sequence, which is for performing despreading on the receiving side, is generated in sync with the detected phase. The DLL carries out control (synchronization tracking) in such a manner that the despreading code sequence on the receiving side will not develop a time shift with respect to a receive signal for which synchronization has been captured even if the receive signal undergoes a change in phase owing to the effects of modulation and noise, etc.  
           [0009]    [0009]FIG. 13 is a diagram illustrating the construction of a DLL circuit  6   b  to which a despreading code generator  6   a  is connected. The despreading code generator  6   a  includes a PN generator  6   a - 1  for generating a despreading code sequence (first PN sequence) A 1 , which is an M sequence. The first PN sequence A 1  is composed of N chips and is generated cyclically at the symbol period T (=N×T C , where T C  represents the chip cycle). The PN generator  6   a - 1  further includes a voltage-controlled oscillator (VCO)  6   a - 2  that is capable of varying the clock frequency (chip frequency) based upon the output of the DLL circuit  6   b . The latter includes a delay circuit  6   b - 1  for delaying the first PN sequence A 1  by one chip cycle and outputting a second PN sequence A 2 ; a despreader (multiplier)  6   b - 2  for multiplying, chip by chip, the first PN sequence A 1  output by the PN generator  6   a - 1  and a receive spread-spectrum data sequence B to thereby effect despreading; a despreader (multiplier)  6   b - 3  for multiplying, chip by chip, the second PN sequence A 2  delayed by one chip and the receive spread-spectrum data sequence B to thereby effect despreading; and adder  6   b - 4  for adding the output of the despreader  6   b - 2  and a signal obtained by inverting the code output by the despreader  6   b - 3 ; and an integrating circuit (low-pass filter)  6   b - 5 .  
           [0010]    The DLL circuit shown in FIG. 13 delays the despreading code sequence A 1  to generate the despreading code sequence A 2  the phase whereof differs by one chip, and uses the despreading code sequences A 1 , A 2  to apply despread processing to the receive data sequence B. However, the DLL circuit can be constructed as shown in FIG. 14. Here the DLL circuit is constructed in such a manner that the receive data sequence B is delayed by a delay circuit  6   b - 1 ′ to generate receive data sequence B′ the phase whereof differs by one chip, and the despreading code sequence A is used to apply despread processing to the receive data sequence B, B′.  
           [0011]    The despreader  6   b - 2  and low-pass filter  6   b - 5  in FIG. 13 function to calculate the correlation between the first PN sequence A 1  and the receive data sequence B. If the phase of the first PN sequence A 1  and the phase of the receive data sequence B match, the maximum output is obtained and, as shown in FIG. 15A, a correlation value R(τ)=1 having the width of one chip cycle is output every symbol. If the phase shifts by one chip cycle or more, the correlation value R(τ) becomes 1/N. The despreader  6   b - 3  and low-pass filter  6   b - 5  function to calculate the correlation between the second PN sequence A 2  delayed by one chip cycle and the receive data sequence B. If the phase of the second PN sequence and the phase of the receive data sequence B match, the maximum output is obtained and a correlation value R(τ) shown in FIG. 15B is output. If the phase shifts by one chip cycle or more, the correlation value R(τ) becomes 1/N. The adder  6   b - 4  adds the output of the despreader  6   b - 2  and a signal obtained by inverting the code output by the despreader  6   b - 3 . As a result, a signal having an S-curve characteristic shown in FIG. 15C with respect to a phase difference T is output via the low-pass filter  6   b - 5 .  
           [0012]    On the basis of the output of the low-pass filter, the voltage-controlled oscillator  6   a - 2  of the despreading code generator  6   a  controls the clock frequency in such a manner that the phase difference T becomes zero. For example, if the phase of the PN sequence (despreading code) leads that of the transmitting-side spreading code contained in the receive data sequence, control is performed so as to make the phase difference zero by lowering the clock frequency. If the phase of the PN sequence (despreading code) lags behind that of the transmitting-side spreading code, control is performed so as to make the phase difference zero by raising the clock frequency.  
           [0013]    The DLL circuit  6   b  in this spread-spectrum system performs despreading at a timing equivalent to the phase difference τ of ±0.5 chip (=±T c /2) with respect to the timing of the desired signal (the spreading code sequence on the transmitting side), obtains the power difference between the signals despread at the respective timings and decides phase advance/delay of the PN sequence (despreading code) based upon the sign (positive or negative) of the power difference, thereby performing path tracking. The timing equivalent to τ=−T c /2 shall be referred to as “early timing” and the timing equivalent to τ=+T c / 2  shall be referred to as “late timing”.  
           [0014]    In the DLL circuits described above, the receive data sequence is described separately for I and Q channels. In actuality, however, the receive data sequence is divided into the I and Q channels and then input to the DLL circuit. FIG. 16 illustrates an example of the construction of a DLL circuit that takes both the I and Q channels into consideration. Components in FIG. 16 identical with those shown in FIG. 12 are designated by like reference characters. The DLL circuit  6   b  includes delay circuits  6   b - 1   i ,  6   b - 1   q  for delaying receive data sequences B I , B Q  of I and Q channels, respectively, by one chip cycle and outputting delayed receive data sequences B I ′, B Q ′; despreaders (multipliers)  6   b - 2   i ,  6   b - 2   q  for multiplying, chip by chip, I- and Q-channel despreading code sequences A I , A Q , which are output by the despreading code generator  6   a , by the receive data sequences B I ′, B Q ′ to thereby effect despreading; and despreaders (multipliers)  6   b - 3   i ,  6   b - 3   q  for multiplying, chip by chip, the I- and Q-channel despreading code sequences A I , A Q  by the delayed receive data sequences B I ′, B Q ′, which are output from the delay circuits, to thereby effect despreading.  
           [0015]    A power calculation unit  6   b - 6  integrates the despread signals from the despreaders  6   b - 2   i ,  2   b - 2   q  over one symbol period, squares the outputs of the integrators and sums the squares to calculate the power value of the despread signals at the early timing. Similarly, a power calculation unit  6   b - 7  integrates the despread signals from the despreaders  6   b - 3   i ,  6   b - 3   q  over one symbol period, squares the outputs of the integrators and sums the squares to calculate the power value of the despread signals at the late timing.  
           [0016]    An adder  6   b - 4  calculates the difference between the power value of the despread signals at the early timing and the power value of the despread signals at the late timing, and an advance/delay decision unit  6   b - 5  instructs the despreading code generator  6   a  to advance/delay the phase of the despreading code sequence based upon an output X from the adder  64 -b. For example, let TH represent a threshold value. If the adder output X is positive and |X|&gt;TH holds, the despreading code generator  6   a  is instructed to advance the phase of the despreading code sequence; if the adder output X is negative and |X|&gt;TH holds, the despreading code generator  6   a  is instructed to delay the phase of the despreading code sequence.  
           [0017]    [0017]FIG. 17 is a diagram expressing FIG. 16 in simplified form. In the description that follows, the DLL circuit will be expressed using this diagram. Furthermore, the despreader  6   b - 2  performs despreading at the early timing and the despreader  6   b - 3  performs despreading at the late timing.  
           [0018]    An example of the DLL will be described for a case where an M sequence of one symbol period is used and the phase difference τ is ±0.5 chip. However, this is not the only arrangement that is possible.  
           [0019]    In a multipath environment, one path interferes with another path if the delay between the paths (the delay time difference between the paths) is too small. As a consequence, the power of a signal despread at whichever of the early and late timings is nearer the timing of the other path becomes too large and results in a DLL control malfunction.  
           [0020]    By way of example, if a path PT 1  at timing  0  in (a) of FIG. 18 is not interfered with by another path, the despread signals at the early timing (=−T c /2) and late timing (=+T c /2) become as shown at (b) and (c) of FIG. 18, respectively, and the S curve becomes zero at time t =0 as indicated at (d) in FIG. 18. Accordingly, if DLL control is carried out so as to eliminate the difference between the despread signal at the early timing and the despread signal at the late timing, synchronization tracking can be achieved at a precision of within one chip. However, if the path PT 1  at timing  0  is interfered with by another nearby path PT 2 , as shown in (a) of FIG. 19, the despread signals at the early timing (=−T c /2) and late timing (=+T c /2) become as shown at (b) and (c) of FIG. 19, respectively. The result is a distorted S curve, which becomes zero at time t=t d  and not at t=0, as indicated at (d) in FIG. 19. Consequently, if DLL control is performed so as to eliminate the difference between the despread signal at the early timing and the despread signal at the late timing, the despreading code sequence will be generated at a timing offset from the original timing by td. The result is a malfunction.  
         SUMMARY OF THE INVENTION  
         [0021]    Accordingly, an object of the present invention is to so arrange it that correct synchronization tracking can be achieved by eliminating the interference component inflicted upon a path of interest by another path in a multipath environment.  
           [0022]    According to the present invention, the foregoing object is attained by a synchronization tracking circuit for synchronizing the phase of a despreading code sequence on a receiving side to the phase of a spreading code sequence on a transmitting side, comprising: a DLL circuit for performing synchronization tracking on a prescribed path of interest among multipaths by DLL control; and an interference-component estimation unit for estimating an interference component inflicted by another path upon the prescribed path of interest in a multipath environment; wherein the DLL circuit eliminates the estimated interference component from a despread signal obtained by despreading a receive signal and controls the phase of the despreading code sequence on the receiving side based upon a signal obtained by elimination of the interference component.  
           [0023]    The interference-component estimation unit estimates the interference component inflicted by the other path upon the path of interest based upon (1) a channel estimation value of the other path, (2) an interpath delay-time difference between the other path and the path of interest, and (3) impulse response of the overall transceiver.  
           [0024]    Thus, in accordance with the synchronization tracking circuit of the present invention, DLL control is performed upon eliminating an interference component that another path inflicts upon a prescribed path of interest in a multipath environment. This makes it possible to achieve correct synchronization tracking.  
           [0025]    Other features and advantages of the present invention will be apparent from the following description taken in conjunction with the accompanying drawings, in which like reference characters designate the same or similar parts throughout the figures thereof. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0026]    [0026]FIG. 1 is a block diagram illustrating an overview of the present invention;  
         [0027]    [0027]FIG. 2 is a waveform diagram of impulse response;  
         [0028]    [0028]FIG. 3 is a block diagram illustrating a first embodiment of a synchronization tracking circuit according to the present invention;  
         [0029]    [0029]FIG. 4 is a diagram useful in describing phase rotation of a pilot symbol;  
         [0030]    [0030]FIG. 5 is a block diagram illustrating a channel estimation unit;  
         [0031]    [0031]FIG. 6 is a block diagram illustrating a second embodiment of a synchronization tracking circuit according to the present invention;  
         [0032]    [0032]FIG. 7 is a diagram useful in describing an interference component;  
         [0033]    [0033]FIG. 8 is a diagram useful in describing the principles of impulse response generation according to another aspect;  
         [0034]    [0034]FIG. 9 is a block diagram illustrating the construction of another impulse-response generating unit;  
         [0035]    [0035]FIG. 10 is a block diagram illustrating a third embodiment of a synchronization tracking circuit according to the present invention;  
         [0036]    [0036]FIG. 11 is a block diagram illustrating a fourth embodiment of a synchronization tracking circuit according to the present invention;  
         [0037]    [0037]FIG. 12 is a block diagram of a CDMA receiver according to the prior art;  
         [0038]    [0038]FIG. 13 is a block diagram of a DLL circuit according to the prior art;  
         [0039]    [0039]FIG. 14 is a block diagram of another example of a DLL circuit according to the prior art;  
         [0040]    [0040]FIGS. 15A, 15B and  15 C are diagrams useful in describing S curves in DLL control according to the prior art;  
         [0041]    [0041]FIG. 16 is a diagram showing the construction of a prior-art DLL circuit that takes I and Q channels into account;  
         [0042]    [0042]FIG. 17 is a block diagram of a prior-art DLL circuit expressed by simplifying FIG. 16;  
         [0043]    [0043]FIG. 18 is a diagram useful in describing S curves in a case where there is no interference between paths; and  
         [0044]    [0044]FIG. 19 is a diagram useful in describing S curves in a case where there is interference between paths. 
     
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0045]    (A) Principles and Overview of the Present Invention  
         [0046]    When a signal that has traversed an ideal propagation path (namely a path of the type along which a transmitted signal is received as is) is filtered by a receive filter, the output signal ν 1 (t) of the receive filter is expressed as follows:  
                 v   1          (   t   )       =       ∑     n   =     -   ∞         +   ∞                         a   n            ∑     k   =   0       N   -   1                         p     k   +   nN            g        [     t   -       (     k   +   nN     )        T       ]                       (   1   )                               
 
         [0047]    where an represents the transmit data (1 or −1), P n  the spreading code (1 or −1), g(t) total impulse response (see FIG. 2) from an input of a transmitter to an output at a receiver, T the chip length (chip cycle) and N a spreading ratio. Further, transmit data a 0  at n=0 is transmit data at the present time, and spreading code sequences at this time p 0 , P 1 , P 2 , . . . , P N−1 .  
         [0048]    If this receive signal is despread by a spreading code (=despreading code sequence) P n , the signal becomes as follows:  
                 v   2          (   t   )       =       1   N            ∑     i   =   0       N   -   1                         p   1          v        (     t   +   lT     )                     (   2   )                   v   2          (   t   )       =       1   N            ∑     n   =     -   ∞         +   ∞                         a   n            ∑     l   =   0       N   -   1                         p   l            ∑     k   =   0       N   -   1                         p     k   +   nN            g        [     t   -       (     k   -   l   +   nN     )        T       ]                             (   3   )                               
 
         [0049]    The signal can be written as follows by splitting it into a desired signal component and an interference component:  
         ν 2 (t)=a 0 g(t)+I(t)  (4)  
         [0050]    The desired signal component, which is the first term on the right side of Equation (4), is that for which  
         n=0, k=ι 
         [0051]    holds in Equation (3). The desired signal component is written as follows:  
                 1   N          a   0            ∑     i   =   0       N   -   1                         p   k   2          g        (   t   )             =         1   N          a   0            ∑     k   =   0       N   -   1                       g        (   t   )           =       a   0          g        (   t   )                   (   5   )                               
 
         [0052]    The interference signal component I(t), which is the second term on the right side of Equation (4), is a signal obtained when the case  
         n=0, k=ι 
         [0053]    is excluded from holding in Equation (3). The interference signal component is written as follows:  
         I(t)= 
         [0054]    [0054]                 1   i          a   0            ∑     i   =   0       N   -   1                         p   l            ∑       k   =   0       k   ≠   l         N   -   1                         p   k          g        [     t   -       (     k   -   l     )        T       ]                 +       1   N            ∑     n   =   0                         a   n            ∑     i   =   0       N   -   1                         p   l            ∑     k   =   0       N   -   1                         p     k   +   nN            g        [     t   -       (     k   -   l   +   nN     )        T       ]                             (   6   )                                 
         [0055]    In I(t), no multiplication takes place between despreading codes of the same number and therefore it is considered that the product of despreading codes takes on a value of 1 or −1 randomly at a 50-50 probability.  
         [0056]    Accordingly, the average power of the interference signal I(t) becomes as follows:  
         P I =&lt;|I(t)| 2 &gt;=(1/N)Σ k≠0 |g(t−kT)| 2   (7)  
         [0057]    so that the power becomes 1/(spreading ratio) of the power that prevailed prior to despreading.  
         [0058]    The main component of an interference signal from another path is the product of the value α i  of the channel (transmission path) and impulse response g(t−Ti) that takes into consideration the interpath delay time T 1  between the path of interest and the other path. The “value” of the channel refers to a quantity that indicates how much attenuation and rotation of phase a signal sustains by transmission along the transmission path. If a despread signal that includes an interference signal from another path is written in the form of a calculation formula, then it can be written as follows from Equation (4):  
               v        (   t   )       =       ∑   i            a   i          [         a   0          g        (     t   -     τ   i       )         +     I   i       ]                 (   8   )                               
 
         [0059]    where g(t) represents the total impulse response of the transceiver, a 0  the transmit data (1 or −1), α i  the value of the channel (channel estimation value) of path i, τ i  the delay time of path i from a path 0 of interest, and Ii the interference component produced when despreading is performed. The interference component Ii is the chip-to-chip interference component produced by the band limitation of a filter or the like and takes on a power of 1/(spreading ratio) owing to despreading, in accordance with Equation (7). The power of the desired signal, however, is assumed to be unchanged by despreading.  
         [0060]    Transforming Equation (8) gives us ν(t)=α 0 [a 0 g(t−τ 0 )+I 0 ]+Σ i≠0 α i [a 0 g(t−τ i )+I i ]  (8)′ 
         [0061]    Accordingly, if despreading is performed at the timing τ 0  of the path  0  (i=0), i.e., at the timing t=τ 0 , then we have  
               v        (     τ   0     )       =         a   0          [         a   0          g        (   0   )         +     I   0       ]       +       ∑     i   ≠   0              a   i          [         a   0          g        (       τ   0     -     τ   i       )         +     I   i       ]                   (   9   )                               
 
         [0062]    where α 0 a 0 g(0) represents the desired signal component. The design is such that the impulse response g(t−τ i ) takes on the maximum value at t=τ i , as shown in FIG. 2, with the amplitude becoming small in terms of average as t departs from τ i . In a multipath environment, therefore, if the spacing between paths is small (i.e., if τ 0 −τ i ) is small, the amplitude of the impulse response g(τ 0 −τ i ) becomes large and path interference from an ith path takes on a large value. FIG. 2 illustrates the interference component g(τ 0 −τ i ) inflicted by path i on path  0 . If the time interval between early timing and late timing in the case of a DLL circuit is represented by T c , the interference from the ith path (path i) will be g(τ 0 −T i −T c /2) and g(T 0 −T i +T c /2) at the early timing and late timing, respectively, and the problem illustrated in FIG. 19 arises as a result.  
         [0063]    Accordingly, the channel estimation value α i  and path timings τ 0 , τ i  are utilized to estimate the interference component Σ i≠0 α i a 0 g(τ 0 −τ i ), and this is subtracted from the despread signal of Equation (8), thereby eliminating the interference component. The channel estimation value a i  is found in the same manner as the channel estimation value used in synchronous detection performed by a CDMA receiver. Further, the path timings τ 0 , τ i  employ the immediately preceding timings (the timings of the immediately preceding symbol) found by the DLL circuit. The impulse response g(t) is a value that is fixed for the particular transceiver. Use is made of a previously measured value or design value and the value is stored in a memory such as a ROM. Thus, a channel estimation value, path timings and impulse response of the overall transceiver are utilized to estimate the interference component, which is then eliminated. As a result, the DLL circuit is allowed to operate based upon the signal solely of the path of interest and it is possible to perform path tracking normally.  
         [0064]    [0064]FIG. 1 is a block diagram illustrating an overview of the present invention. Here a DLL circuit  11  controls the phase of a despreading code sequence by DLL control and includes an interference eliminating unit  11   e . A despreading code sequence generator  21  generates a despreading code sequence at a timing instructed by a searcher (matched filter), not shown, and advances or delays the phase of the despreading code sequence in accordance with a phase advance/delay command from the DLL circuit  11 . An interference signal generator  12  estimates the interference component Σ i≠0 α i a 0 g(τ 0 −τ i ) using the channel estimation value α i , path timings τ 0 , τ i  and impulse response g(t) (FIG. 2) of the overall transceiver, and inputs this component to the DLL circuit  11 . The interference eliminating unit  11   e  eliminates the interference component from the despread signal obtained by despreading the receive signal, and the DLL circuit  11  eliminates the interference component, as a result of which DLL control can be carried out based upon the signal solely of the path of interest. This makes normal path tracking possible.  
         [0065]    (B) Embodiments  
         [0066]    (a) First Embodiment  
         [0067]    [0067]FIG. 3 is a block diagram illustrating a first embodiment of a synchronization tracking circuit according to the present invention. The DLL circuit is expressed in simplified form in a manner similar to that of FIG. 17. Accordingly, the signals of various components up to input to a power calculation section in the DLL circuit are complex signals.  
         [0068]    The signal received by the radio receiving unit of a CMDA receiver is converted to digital data by an A/D converter  10 , and the digital data generated by the conversion is input as a receive data sequence B to the DLL circuit  11  of the path of interest. The receive data sequence B corresponds to spread data obtained by spreading transmit data by a spreading code on the transmitting side. The interference signal generator  12  estimates an interference component that is inflicted upon a prescribed path of interest by another path in a multipath environment and inputs this interference component to the DLL circuit  11 . More specifically, the interference signal generator  12  estimates the interference component inflicted by the other path upon the path of interest based upon (1) a channel estimation value of the other path, (2) a delay time difference between the other path and the path of interest, and (3) impulse response of the overall transceiver. The DLL circuit  11  eliminates the interference component, which is inflicted by the other path, from the despread signal obtained by despreading the receive data sequence, and outputs a signal PCS for controlling the phase of the despreading code sequence on the receiving side based upon the signal obtained. A channel estimation unit  13  uses a data sequence B″, which is obtained by delaying the receive data by T c /2, to obtain a channel estimation value through a method similar to that used in channel estimation in synchronous detection, and inputs the channel estimation value to the interference signal generator of the other path.  
         [0069]    The DLL circuit  11  includes delay circuits  11   a ,  11   b  for delaying the receive data sequence B by T c /2 (where T c  represents the chip cycle) at a time, a first despreader  11   c  for despreading the receive data sequence B using a despreading code sequence, and a second despreader  11   d  for despreading a receive data sequence B′, which has been delayed by a total delay of T c , using a despreading code sequence. If timing that follows the timing of the receive data sequence B by the time T c /2 is regarded as the timing of the spreading code on the transmitting side, then the first despreader  11   c  despreads the receive data at a timing (early timing) advanced in phase by T c /2 relative to the timing of the spreading code sequence on the transmitting side, and the second despreader  11   d  despreads the receive data at a timing (late timing) delayed in phase by T c /2 relative to the timing of the spreading code sequence on the transmitting side.  
         [0070]    The DLL circuit  11  further includes the interference eliminating unit  11   e  for eliminating interference received from another path by subtracting an interference component IS from the despread signal of the early timing, and an interference eliminating unit  11   f  for eliminating interference received from the other path by subtracting an interference component IS′ from the despread signal of the late timing. The interference signals IS, IS′ both are interference signals which other the other path inflicts upon the path of interest but the values of these signals differ. The reason for this is as follows: If we let τ 0  represent the timing (path timing) of a desired signal on the path of interest, the first despreader  11   c  despreads the receive data at the timing (early timing) of (τ 0 −T c /2), and the second despreader  11   d  despreads the receive data at the timing (late timing) of (τ 0 +T c /2). Accordingly, if we let Ti represent the path timing of another path, the delay time from the path timing of the other path to the early timing of the first despreader  11   c  will be (τ i −τ 0 +T c /2), and the delay time from the path timing of the other path to the late timing of the second despreader  11   d  will be (τ i −τ 0 −T c /2). Thus the delay times are different. As a consequence, the interference signals IS, IS′ that depend upon the delay time difference between the other path and the path of interest have different values.  
         [0071]    The DLL circuit  11  further includes a power calculation unit  11   g  for calculating the power of the signal obtained by eliminating the interference signal IS from the despread signal of the early timing, a power calculation unit  11   h  for calculating the power of the signal obtained by eliminating the interference signal IS′ from the despread signal of the late timing, an arithmetic unit  11   i  for calculating the power difference, and a sign discrimination unit  11   j  for outputting a phase control signal PCS that controls the phase of the despreading code sequence on the receiving side based upon the power difference. If the sign discrimination unit  11   j  discriminates a positive sign, it outputs a phase control signal PCS that retards the phase of the despreading code sequence on the receiving side in such a manner that the power difference will become zero; if the sign discrimination unit  11   j  discriminates a negative sign, it outputs a phase control signal PCS that advances the phase of the despreading code sequence on the receiving side in such a manner that the power difference will become zero.  
         [0072]    The channel estimation unit  13  obtains a channel estimation value through a method similar to that used for channel estimation is synchronous detection. In CDMA communication, a pilot symbol P undergoes rotation of phase owing to transmission. If a signal point position vector P A  (see FIG. 4) of this signal is known on the receiving side, then the phase rotation angle θ and amplitude attenuation of the symbol resulting from transmission can be obtained because an ideal signal point position vector PIDL of the pilot symbol is already known. The phase rotation angle θ and attenuation become the channel estimation values. FIG. 5 is a block diagram illustrating the channel estimation unit  13 . The latter includes despreaders  13   a ,  13   b  for respectively despreading I- and Q-channel sequences B I ″, B Q ″ of a data sequence, which is obtained by delaying the receive data sequence B by T c /2, by I- and Q-channel despreading code sequences on the receiving side. Switches  13   c ,  13   d  are closed by at the pilot receive timing, whereby I- and Q-channel components Ip, Qp of the pilot symbol are input to a channel estimation value calculation unit  13   e . Whenever the I- and Q-channel components Ip, Qp of the pilot symbol enter, the channel estimation value calculation unit  13   e  uses these signals and I- and Q-channel components Ikp, Qkp of an already known pilot symbol to calculate I- and Q-channel components of the channel estimation signal. Integrators  13   f  and  13   g  average the I- and Q-channel components, respectively, and output channel estimation values It and Qt, respectively.  
         [0073]    The interference signal generator  12  generates the interference components IS, IS′ inflicted upon a path PTO of interest by another path PT i (i=1, 2, . . . ). The interference components inflicted upon a path PTO of interest by another path PT i  are α i a 0 g(τ 0 −τ i ), as indicated by Equation (9), where a 0  represents the transmit data (1 or −1), α i  the channel estimation value (=Iti+jQti) of path PT i , τ 0  the path timing of the path PTO of interest, τ i (i=1, 2, . . . ) the path timing of path PT i , (τ c −τ i ) the delay time from path PTO to path PTi, and g(t) the impulse response.  
         [0074]    An impulse response generator  12   a  stores the correspondence between times and impulse response values (see FIG. 2) discretely in a storage unit such as a ROM or RAM, reads an impulse response value g(τ 0 −τ i ), which conforms to a delay time difference (τ 0 −τ i ) requested from an interference signal estimation unit  12   b , out of the storage unit and outputs this impulse response value. The interference signal estimation unit  12   b  uses channel estimation values α i (i=1, 2, . . . ), path timing τ i  and impulse response value g(t), which enter from other fingers, to estimate the interference signal IS of the early timing and the interference signal IS′ of the late timing, and outputs these interference signals.  
         [0075]    The interference signal IS is estimated in accordance with  
         IS=Σ i α 0 a 0 g(τ 0 −τ c /2)i=1, 2, . . .   (10)  
         [0076]    and the interference signal IS′ is estimated in accordance with  
         IS′=Σ i α i a 0 g(τ 0 −τ i −T c /2)i=1, 2, . . .   (10)′ 
         [0077]    Thus, in accordance with the first embodiment, the interference component which another path inflicts upon a path of interest is estimated by utilizing a channel estimation value α i , path timing τ i  and impulse response value g(t) of the overall transceiver, and DLL control is carried out upon eliminating this interference component from the receive signal. As a result, DLL control can be carried out based upon a signal solely of a path of interest. This makes normal path tracking possible.  
         [0078]    (b) Second Embodiment  
         [0079]    [0079]FIG. 6 is a block diagram illustrating a first embodiment of a synchronization tracking circuit according to the present invention. The second embodiment illustrates an example of a case where a single path interferes with a path of interest. Components shown in FIG. 6 identical with those of the first embodiment of FIG. 3 are designated by like reference characters.  
         [0080]    Here the interference signal estimation unit  12   b includes a first impulse-response calculation unit  21  for calculating and outputting an impulse response value g(T   1 −T 0 +T c /2) at the early timing (T 1 −T 0 +T c /2), and a second impulse-response calculation unit  22  for calculating and outputting an impulse response value g(T 1 −T 0 −T c /2) at the late timing (T 1 −T 0 −T c /2).  
         [0081]    The interference signal estimation unit  12   b further includes a first multiplier  23  for multiplying the impulse response value g(T   1 −T 0 +T c /2) by the channel estimation value a 1  of path PT 1  to produce the interference signal IS expressed by  
         IS=α 1 g(T 1 −T 0 +T c   /2)    
         [0082]    This signal is input to the interference eliminating unit  11   e . The interference signal estimation unit  12   b further includes a second multiplier  24  for multiplying the impulse response value g(T   1 −T 0 −T c /2) by the channel estimation value α 1  of path PT 1  to produce the interference signal IS′ expressed by  
         IS′=α 1 g(T 1 −T 0 −T c /2)  
         [0083]    This signal is input to the interference eliminating unit  11   f . The interference eliminating unit  11   e  eliminates the interference from the other path by subtracting the interference signal IS from the despread signal of the early timing, and the interference eliminating unit  11   f  eliminates the interference from the other path by subtracting the interference signal IS′ from the despread signal of the late timing. This is followed by performing DLL control that is similar to that of the first embodiment.  
         [0084]    The second embodiment is such that if there is one path that interferes with the path of interest, the DLL circuit of the path PT 0  of interest performs DLL control using a signal from which interference inflicted by the interfering path PT 1  has been eliminated. This makes it possible to perform correct synchronization tracking control.  
         [0085]    [0085]FIG. 7 is a diagram useful in describing an interference component and illustrates the impulse response of path PT 1 . Here T 1  represents the despreading timing of path PT 1  the angle of phase rotation on path PT 1 and θ   1  the attenuation [channel estimation value α 1 =A 1 exp(jθ 1 )] on this path, and To the timing of despreading on path PT 0 . The interference path PT 1  inflicts upon the path PT 0  of interest is the impulse response value at timing T 0  . From FIG. 7, this is  
         A 1 exp(jθ 1 )g(T 1 −T 0 )  
         [0086]    The interference signals I s , I s &#39; at the early timing (T 0 −T c /2) and late timing (T 0 +T c /2), respectively, are as follows:  
         [0087]    IS=A 1 exp(jθ 1 )g(T 1 −T 0 +T c /2)  
         [0088]    IS′ =A 1 exp(jθ 1 )g(T 1 −T 0 −T c /2)  
         [0089]    (c) Alternative Construction of Impulse-response Generator  
         [0090]    [0090]FIG. 8 is a diagram useful in describing the principles of impulse response generation according to another aspect, and FIG. 9 is a diagram illustrating an another construction of the impulse response generator. In the first and second embodiments, the impulse response generator  12   a  stores the correspondence between times and impulse response values discretely in a storage unit such as a ROM or RAM, reads an impulse response value g(T 1 −T 0 ±T c /2), which conforms to the delay time difference (T 1 −T 0 ±T c /2), out of the storage unit, and outputs the impulse response value. According to such an implementation, however, a large-capacity memory is necessary to store the impulse response values.  
         [0091]    Accordingly, the impulse response value at a predetermined time shown in FIG. 8 is approximated by ½ n  of the peak value (where n is a positive integer), and the correspondence between time and n is stored in a storage unit (a bit-shift quantity storage unit)  31  discretely. Further, the peak value I peak  of the impulse response is stored in a storage unit  32 . An impulse-response calculation unit  33  obtains the n that corresponds to the delay time difference (T 1 −T 0 ±T c /2), shifts the peak value I peak  by n bits, calculates the impulse response value g(T 1 −T 0 ±T c /2) and outputs the value. If this arrangement is adopted, the memory capacity required can be reduced.  
         [0092]    (d) Third Embodiment  
         [0093]    If the delay between paths in a multipath environment is large, the effect of other paths upon a path of interest is small. If the delay between paths is small, however, other paths do have a large influence upon the path of interest. In the third embodiment, therefore, the interference component is estimated and eliminated only in a case where the delay between paths is less than a threshold value.  
         [0094]    [0094]FIG. 10 is a block diagram illustrating a third embodiment of a synchronization tracking circuit according to the present invention. Components shown in FIG. 10 identical with those of the second embodiment of FIG. 6 are designated by like reference characters. This embodiment differs from the second embodiment in the following respects:  
         [0095]    (1) A path spacing monitoring unit  25  is provided. This unit obtains the time difference (interpath delay-time difference) between each of the early and late timings of the path PTO of interest and the path timing of path PT 1  compares the interpath delay-time difference with a set time T s  and outputs switch open/close signals SOC1, SOC2 based upon the comparison.  
         [0096]    (2) Switches  26 ,  27 , which are opened/closed by the switch open/close signals SOC1, SOC2, respectively, are provided.  
         [0097]    More specifically, the path spacing monitoring unit  25  (1) outputs the switch open/close signal SCO1 the logic level whereof is high when the interval (=T 1 −T 0 +T c /2) between the early timing of the path PTO of interest and the path timing of the path PT 1  is equal to or less than the set time T s , and (2) outputs the switch open/close signal SCO1 the logic level whereof is low when the interval(=T 1 −T 0 +T c /2) is greater than the set time T s . Further, the path spacing monitoring unit  25  (1) outputs the switch open/close signal SCO2 the logic level whereof is high when the interval (=T 1 −T 0−T   c /2) between the late timing of the path PTO of interest and the path timing of the path PT 1  is equal to or less than the set time T s , and (2) outputs the switch open/close signal SCO2 the logic level whereof is low when the interval(=T 1 −T 0 −T c /2) is greater than the set time T s .  
         [0098]    The switch  26  (1) closes when the switch open/close signal SCO1 is at the high level (=T 1 −T 0 +T c /2≦T s ), thereby inputting the impulse response value g(T 1 −T 0 +T c /2) to a multiplier  23 , and (2) opens when the switch open/close signal SCO1 is at the low level (=T 1 −T 0 +T c /2&gt;T s ), thereby inputting zero to a multiplier  23 . Further, the switch  27  (1) closes when the switch open/close signal SCO2 is at the high level (=T 1 −T 0 −T c /2≦T s ), thereby inputting the impulse response value g(T 1 −T 0 −T c /2) to a multiplier  24 , and (2) opens when the switch open/close signal SCO2 is at the low level (=T 1 −T 0 −T c /2&gt;T s ), thereby inputting zero to a multiplier  24 .  
         [0099]    Thus, if the delay-time difference between paths is small and, hence, the switch open/close signal SCO1 is at the high level (i.e., when T 1 −T 0 +T c /2&lt;T s  holds), the multiplier  23  inputs the interference signal Is=α 1 g(T 1 −T 0 +T c /2) to the DLL circuit  11 , and the latter performs DLL control upon eliminating the interference component IS. However, if the delay-time difference between paths is large and, hence, the switch open/close signal SCO1 is at the low level (i.e., when T 1 −T 0 +T c /2&gt;T s  holds), the multiplier  23  outputs an interference signal IS that is equal to zero. As a result, the DLL circuit  11  performs DLL control without eliminating the interference component.  
         [0100]    Further, if the delay-time difference between paths is small and, hence, the switch open/close signal SCO2 is at the high level (i.e., when T 1 −T 0 −T c /2&lt;T s  holds), the multiplier  24  inputs the interference signal IS′=α 1 g(T 1 −T 0 −T c /2) to the DLL circuit  11 , and the latter performs DLL control upon eliminating the interference component IS′. However, if the delay-time difference between paths is large and, hence, the switch open/close signal SCO2 is at the low level (i.e., when T 1 −T 0 −T c /2&gt;T s  holds), the multiplier  24  outputs an interference signal IS′ that is equal to zero. As a result, the DLL circuit  11  performs DLL control without eliminating the interference component.  
         [0101]    The foregoing is for a case where there is one path that interferes with the path of interest. However, control can be performed in a similar manner also in cases where multiple path interfere with the path of interest.  
         [0102]    In accordance with the third embodiment, therefore, an interference component is eliminated only if the interference component is large, thereby making it possible to perform correct synchronization tracking.  
         [0103]    (e) Fourth Embodiment  
         [0104]    In a multipath environment, the phases of delayed waves differ from one another. Accordingly, highly precise synchronization tracking can be achieved if DLL control is performed so as to find the difference between signals obtained eliminating an interference signal from each of the despread signals of both the early and late timings, rotating phase using a channel estimation value and deciding advance/delay based upon the sign (positive or negative) of this signal.  
         [0105]    [0105]FIG. 11 is a block diagram illustrating a fourth embodiment of such a synchronization tracking circuit. Components shown in FIG. 11 identical with those of the second embodiment of FIG. 6 are designated by like reference characters. This embodiment differs from the second embodiment in that the phase of a despread signal from which an interference signal has been eliminated is rotated using a channel estimation value and the phase control signal PCS is output in accordance with the sign of the signal obtained by phase rotation.  
         [0106]    An arithmetic unit  11   m  obtains, by complex calculation, the difference between despread signals of the early and late timings from which an interference signal has been eliminated by the interference eliminating units  11   e ,  11   f , and outputs a complex signal representing this difference. A phase rotation unit  11   n  uses the channel estimation values It, Qt, which are output by the channel estimation unit  13 , to rotate the phase of the complex signal output from the arithmetic unit  11   m . The sign discrimination unit  11   j  outputs the phase control signal PCS based upon the sign of the I component of the complex signal obtained by phase rotation.  
         [0107]    In accordance with the fourth embodiment, phase rotation is performed using a channel estimation signal and phase advance/delay is judged based upon the signal of the signal obtained by this phase rotation. This makes possible highly precise synchronization tracking.  
         [0108]    Thus, in accordance with the present invention, DLL control is performed upon eliminating an interference component that another path inflicts upon a path of interest in a multipath environment. As a result, correct synchronization tracking can be carried out.  
         [0109]    Further, in accordance with the present invention, an interference component is estimated correctly based upon (1) a channel estimation value of the other path, (2) a path-to-path delay-time difference between the other path and the path of interest, and (3) impulse response of the overall transceiver. The effects of this interference component can be eliminated.  
         [0110]    Further, in accordance with the present invention, the impulse response value is approximated by ½ n  of the peak value (where n is a positive integer), the correspondence between time and n is stored discretely and the impulse response value is calculated upon shifting the peak value by n bits. As a result, the required memory capacity of the impulse response generator can be reduced.  
         [0111]    Further, in accordance with the present invention, phase is rotated using a channel estimation value and phase advance/delay is discriminated based upon the sign of the signal obtained by phase rotation. This makes it possible to perform highly precise synchronization tracking.  
         [0112]    As many apparently widely different embodiments of the present invention can be made without departing from the spirit and scope thereof, it is to be understood that the invention is not limited to the specific embodiments thereof except as defined in the appended claims.