Abstract:
A radio receiver processing path has a mixer with active interference/blocker cancellation to reduce the intensity of leaked and undesired signals by using a replica of the transmitted signal, emulating the phase and attenuation through the leakage path and subtracting the emulated signal within the mixer. Intermodulation distortions are predicted through the use of nonlinear modeling in the digital baseband between the baseband transmitter and baseband receiver and subsequently subtracted from the received signal. The nonlinear basis functions are combined to model the composite nonlinearity in the signal path based on digital baseband transmitted data. The modeled nonlinearity is subtracted from the received signal, and the result is observed and used to guide the nonlinear modeling parameters using self-contained control loops.

Description:
CROSS REFERENCES TO RELATED APPLICATIONS 
     This application claims priority from U.S. Provisional Patent Application Ser. No. 61/726,523, entitled “Method and System for Mitigating the Effects of a Transmitted Blocker and Distortions Therefrom in a Radio Receiver”, filed on Nov. 14, 2012, which is hereby incorporated by reference as if set forth in full in this application for all purposes. 
    
    
     FIELD OF THE INVENTION 
     The present invention relates to the field of radio communication systems and methods and systems for improving the quality of received signals by reducing the portion of other signals that are leaked into the receive signal path and/or removing the associated intermodulation products that fall into or near the receive band. 
     BACKGROUND 
     A conventional frequency division duplex (FDD) radio system simultaneously transmits a signal on one frequency band while receiving another signal on another frequency band. The context is illustrated in  FIG. 2 a -2 b    for communication between two radios. 
       FIG. 2 a    illustrates two radios: Radio-A  210  and Radio-B  230 . Radios  210  and  230  each contain an associated transmitter  211 ,  231  and receiver  212 ,  232 , respectively. If Radio-A  210  is upstream or closer to the central office station than Radio-B  230 , then the signal transmitted from Radio-A  210  and received by Radio-B  230  is called the downlink (DL) signal  220 . Similarly, in such a case, the signal transmitted from Radio-B  230  and received by Radio-A  210  is called the uplink (UL) signal  240 . One illustrative context includes Radio-A  210  in a base station and Radio-B  230  in a mobile device, such as a mobile phone. 
       FIG. 2 b    illustrates key aspects of a frequency spectrum  250  in an FDD system such as drawn in  FIG. 2 a   . The DL signal band  260  and UL signal band  280  occupy disjoint but not necessarily contiguous frequency bands.  FIG. 2 b    illustrates the DL band  260  as being lower in frequency than the UL band  280 , but those skilled in the art understand that that need not be the case in general. 
     Generally, there is also an inactive or unused portion of the spectrum between the DL band  260  and UL band  280  called the guard band (GB)  270 . Because radio signals cannot be perfectly confined to a frequency band, i.e. they leak energy outside their prescribed band, the GB  270  serves to reduce the impact of leakage as the strongest leakage is typically closest to the signal&#39;s main band. However, as is known in the art, even with the use of a GB, the amount of isolation is finite and some degree of leakage always exists. 
     The basic elements of a radio  100  are shown in  FIG. 1  and may apply to either Radio-A  210  or Radio-B  230  of  FIG. 2A . 
     For the sake of clarity and without loss of generality, the remainder of the document will describe all radios as if they were the upstream Radio-A  210  that is transmitting a DL signal  220  and receiving an uplink signal  240 , but it is obvious to those skilled in the art that the description could be equally applied to downstream Radio-B  230 . 
     In the example of  FIG. 1 , the digital data is encoded into a baseband or low intermediate-frequency (IF) waveform and output in digital form from the baseband transmitter (BB Tx)  110  to a digital-to-analog converter (DAC)  120  which converts the digital signal into a continuous-time analog baseband or low IF waveform. The output of DAC  120  is then modulated up in frequency or up-converted by an up-converting mixer  130 . The up-converting mixer  130  modulates the baseband output of DAC  120  up to the carrier frequency in the DL band  260  as set by local oscillator  135 . The radio-frequency (RF) output of the up-converting mixer  130  is then amplified by one or more amplifiers including a dedicated power amplifier (PA)  140 . The PA output drives one portion  151  of a duplexer  150  to an antenna  155 . Filter  151  in the duplexer  150  serves to attenuate transmitted signal components outside the desired transmit or DL band  260  while maintaining the signal strength within the transmit or DL band  260 . The antenna  155  radiates the transmit or DL signal  220  electromagnetically to another radio. 
     In the example of  FIG. 1 , the receiver portion of radio  100  works in a similar manner but in reverse. In particular, antenna  155  receives a radio signal  240  in the receive or UL band  280  and filters that with a filter  152  in the duplexer  150 . Filter  152  in the duplexer  150  serves to attenuate signal components outside the receive or UL band  280  while preserving the strength of the signal within the receive or UL band  280 . The output of filter  152  in the duplexer  150  is fed to a low-noise amplifier (LNA)  160  to amplify the signal. The output of LNA  160  is then down-converted from the receive or UL band  280  to baseband by the down-converting mixer  170  where the amount of frequency down-conversion is controlled by the frequency of local oscillator  175 . The low-IF or baseband signal output by the down-converting mixer  170  is quantized by the receiver analog-to-digital converter  180  to form a digital representation of the baseband or low-IF signal which is then passed to the baseband receiver (BB Rx)  190  for demodulation and decoding. 
     Imperfect isolation in the duplexer filters  151 ,  152  and nonlinear characteristics of various radio components such as but not limited to the up-converting mixer  130 , PA  140 , duplexer  150 , antenna  155 , LNA  160 , down-converting mixer  170 , and ADC  180  can lead to the transmitted DL signal corrupting or reducing the signal fidelity of the received UL signal. Some of these impairments will be described with the aid of  FIGS. 3-6  illustrating the signal spectrum at various points in the signal path. 
       FIG. 3  illustrates a signal spectrum  300  of the transmitted signal such as may be observed at the output of the PA  140 . 
     In the example transmitted DL band  260 , there reside a plurality (four are drawn but there may be more or fewer) of channels  310   a - d  carrying data at a prescribed transmit power level  320 . An undesired, but practically unavoidable, artifact of the transmitted channels  310  are intermodulation (IM) products  340 . The third order IM products  340   a  and  340   b  are closest in frequency to the DL band  260 , followed by the fifth order IM products  340   c  and  340   d , then the seventh order IM products  340   e  and  340   f , and so on. The received UL band  280  will often overlap with the frequency of one or more of these IM products, and hence, the received UL signal  240  is subject to being distorted by these IM products. For that reason, communications standards such as CDMA, GSM, LTE, and others, will specify a minimum adjacent channel leakage ratio (ACLR) defining the ratio of the power  320  of the intended transmit DL signal to the maximum power  330  from any of the IM products  340  to limit the amount of distortion. 
     If the spectrum  300  in  FIG. 3  were that of the PA  140  output, then the effect of the filter  151  in the duplexer  150  is illustrated in  FIG. 4  showing the spectrum of the resulting transmit DL signal at the antenna  155 . 
     In this example, the duplexer transmit filter  151  has little effect on the intended transmit signal  310  in the transmit DL band  260 , but signal components outside the transmit DL band  260  are attenuated. Consequently, the IM products  340  have now been attenuated to lower levels shown by the smaller IM products  440   a - f.    
     Spectrum  500  in  FIG. 5  similarly illustrates the signal spectrum of the received signal at the output of the duplexer receive filter  152 . 
     In this example, the duplexer receiver filter  152  is configured to pass signals in the received UL band  280  unabated. Thus, the offending IM product in this band (drawn as the fifth IM product  440   d ) is unaffected along with the desired received signal  550  received off the air from the antenna  155 . Other signal components outside the received UL band  280  have been attenuated in this example, such as the transmitted DL signal  510  and the other IM products  540   a - c, e, f.    
     Spectrum  600  in  FIG. 6  illustrates the example signal as it may be after the LNA  160  or down-converting mixer  170 , or ADC  180 . 
     In this example, nonlinear behavior in any of these elements will tend to increase the levels of the transmitted signal IM products  640   a - f  due to intermodulation since the transmitted signal  510  is typically the strongest signal component even after all filtering in the duplexer  150 . Of particular concern is that the level of the IM product in the receive UL band  280  may increase, and consequently, degrade the signal to interference ratio (a measure of signal fidelity) of the received signal  550  to the interferer  640   d.    
     As is well understood by those skilled in the art, radio systems (especially consumer mobile devices) are increasing their data capacity and consequently demanding more stringent signal fidelity. A common limitation in radio systems is the fidelity of the received signal from a distant device and, in particular, its corruption by leaked interference as described above. The present invention addresses this interference in multiple means to (i) reduce the leaked transmit blocker  510   a - d  in the receive path and (ii) reduce the offending IM product(s)  640  in the received UL band  280  from nonlinear components in the transmit and receive paths. 
     SUMMARY 
     Embodiments of the invention reduce the deleterious effect of a leaked transmit signal on a received radio signal and improve the signal fidelity of a received radio signal by employing a canceling down-mixer in the receive path prior to sampling an analog receive path signal and/or an adjacent channel leakage cancellation block to remove leakage noise from a digitized receive path signal. 
     In an embodiment, a transmit path signal is sensed and down-converted. The down-converted transmit path signal is scaled and processed, and then subtracted from a an analog receive path signal using a canceling down-mixer. The resulting signal is then digitized for further processing, including demodulation and optionally adjacent channel leakage cancellation. 
     In an embodiment, intermodulation noise is predicted using nonlinear modeling in the digital baseband between the baseband transmitter and baseband receiver. For example, non-linear basis functions may be combined to model the composite nonlinearity in the signal path based on digital baseband transmitted data. The modeled approximation of intermodulation noise is subsequently subtracted from the receive path signal. A further embodiment may use adaptive modeling of the intermodulation noise to guide the nonlinear modeling parameters using self-contained control loops. 
     A further understanding of the nature and the advantages of particular embodiments disclosed herein may be realized by reference of the remaining portions of the specification and the attached drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The invention will be described with reference to the drawings, in which: 
         FIG. 1  illustrates a conventional FDD radio; 
         FIGS. 2 a -2 b    illustrate a system in which two frequency-division duplex (FDD) radios communicate using an example frequency spectrum including a guard band; 
         FIGS. 3-6  illustrate example signal spectrums showing various sources of noise and distortion; 
         FIGS. 7 a -7 b    illustrate radio systems including canceling down-converting mixers according to embodiments of the invention; 
         FIGS. 8 a -8 c    illustrate example canceling down-converting mixers according to embodiments of the invention; 
         FIG. 9  illustrates an example cancellation block according to an embodiment of the invention; 
         FIG. 10  illustrates a radio utilizing digital pre-distortion in the transmit path and suitable for use with embodiments of the invention; and 
         FIGS. 11 a  and 11 b    illustrate radio systems including canceling down-converting mixers and adjacent channel leakage cancellation according to embodiments of the invention. 
     
    
    
     DETAILED DESCRIPTION OF EMBODIMENTS 
     Embodiments of the invention may employ one or more of the following corrective aspects to reduce the deleterious effect of a leaked transmit signal on a received radio signal and improve the signal fidelity of a received radio signal. 
       FIG. 7 a    illustrates a radio system  700   a  that reduces leakage and intermodulation noise according to an embodiment of the invention. System  700   a  is similar to that of system  100  in  FIG. 1 . 
     However, system  700   a  replaces the conventional down-converting mixer  170  of  FIG. 1  with a cancelling down-converting mixer (CDM)  770 . As shown in  FIG. 7 a   , system  700   a  directs a copy of an analog transmit signal at the output of DAC  120  through filter  725  to the CDM  770 . The CDM  770  serves to replicate the leaked DL blocker signal  510   a - d  arising from the transmitted DL signal  310  as the latter goes through the duplexer  150  and to subtract the DL blocker signal from the receive path signal to improve the signal fidelity of the received UL signal  240 . In this embodiment, filter  725  modifies the analog copy of the transmit signal to create an approximation of the leakage noise introduced into the radio receive path by the radio transmit path. The filter  725  could include an amplifier and/or one or more delay elements to implement functionality ranging from a simple amplifier to a phase delay and up to a pole-zero filter as known to those skilled in the art. The output of filter  725  is then sent to the CDM  770 . The CDM  770  reduces the leakage noise included in a receive signal by subtracting the filter&#39;s  725  approximation of leakage noise from the receive path signal. 
     In an alternate embodiment, a digital version of the transmit signal may be processed in the digital domain to construct an approximation of the actual leakage noise.  FIG. 7 b    illustrates a radio system  700   b  that reduces leakage and intermodulation noise according to an embodiment of the invention. System  700   b  is similar to that of system  700   a  in  FIG. 7 a   . However, system  700   b  directs a digital version of the transmit signal to digital processing block  771 . Like filter  725  in system  700   a , digital processing block  771  uses the transmit signal to construct an approximation of the leakage noise introduced into the receive path. Digital processing block  771  may utilize digital multiplications (for amplification or attenuation) and/or digital delays to implement filtering operations as simple as signal scaling or linear phase adjustment to more complex filtering operations such as finite impulse response (FIR) and/or infinite impulse response (IIR) filters as known to those skilled in the art. DAC  772  converts the leakage approximation to the analog domain, suitable for use in CDM  770 . 
     A second change in systems  700   a  and  700   b  from system  100  is the addition of the adjacent channel leakage canceler (ACL) block  785  after the ADC  180 . The ACL canceler block  785  serves to replicate the nonlinear intermodulation (IM) products  640   a - f  arising from the transmitted DL signal  310  as the latter goes through various nonlinear components and to subtract the replicated IM products from the receive path signal to improve the signal fidelity of the received UL signal  240 . 
       FIG. 8 a    and  FIG. 8 b    show two embodiments of the canceling down-converting mixer,  770   a  and  770   b , used by systems  700   a  and  700   b . In the embodiment  770   a  illustrated in  FIG. 8 a   , the leakage approximation signal  803  is up-converted by up-converting mixer/multiplier  820 , controlled by local oscillator  135  which may optionally be the same as local oscillator  175 . Subtractor  830  subtracts the resulting signal  805  in the RF domain from the RF receive signal  801 , which may be an output signal of LNA  160 . CDM  770   a  filters the results with filter  840  and then down-converts the filtered signal to intermediate frequency or baseband by mixer/multiplier  810 , which is also driven by local oscillator  175 . In alternative embodiments, filter  840  may be positioned before subtracter  830  or it may be omitted. An advantage of the embodiments  770   a  is that the leaked transmit signal in DL band  260  is reduced prior to mixer/multiplier  810 , which due to its nonlinear nature is a potential generator of IM noise. 
       FIG. 8 b    shows a simpler configuration  770   b  of the canceling down-converting mixer. CDM  770   b  receives a receive path signal  801 . Mixer  810  mixes the receive path signal  801  with a local oscillator signal  175  to create a baseband or IF receive path signal. Optional filter  840  then processes the down-mixed receive path signal. CDM  770   b  then subtracts a leakage approximation signal  803  from this receive path signal. 
     The CDMs  770   a  and  770   b  both have the advantage that their output signal  804  has a significantly reduced transmit blocker from DL channel  260 , significantly enhancing the dynamic range of the desired receive signal at the input of analog-to-digital converter  180 , thereby much reducing the requirements and cost for this ADC. 
     Embodiments of the CDM may further include aspects of a bandpass mixer, such as from U.S. Pat. No. 8,798,570 B2. A bandpass mixer is an architecture for a mixer that provides an integrated filter function, which for the purposes of this document may be a bandpass filter or any other type of filter. The characteristics of the integrated filter are transformed from a band of low frequencies, for instance baseband or intermediate frequencies, to a band of RF frequencies such as seen for desired and undesired input signals of the bandpass mixer architecture. This transformation may be a simple frequency shift, such that steep filter characteristics that are available easily at low frequencies now become available at RF frequencies. If the frequency shift is obtained through the use of a multiplier and local oscillator, then the RF filter characteristics of the bandpass mixer can be tightly controlled. If the same local oscillator is used for down-converting the bandpass mixer input signal and up-converting (shifting) the filter characteristics, then the bandpass mixer can achieve accurately tracking narrowband RF filtering. 
     Combining the aspects of a canceling down-converting mixer and a bandpass mixer will make a canceling bandpass mixer, or CBP, an embodiment  770   c  of which is shown in  FIG. 8 c   . The CBP, for instance by utilizing a sharply frequency-dependent input impedance, allows for reducing the leaked transmit signal  510 , without impacting the desired receive signal  550 . This could for instance be a controlled frequency-dependent input impedance for gm-type LNAs that output current (rather than voltage) as a signal. By presenting a nominal load at frequencies in the received UL band  280 , these signal components (e.g. received UL signal  550 ) pass unimpeded. But by also presenting a significantly lower impedance closer to 0 ohms at frequencies outside the received UL band  280 , these signal components (e.g. the leaked transmit blocker  510   a - d ) are significantly attenuated. 
     The receiver local-oscillator  175  can be used to up-convert this reconstructed transmit signal up to RF within the CBP mixer. This is illustrated in  FIG. 8 c   . In an embodiment, CBP mixer system  770   c  shows transconductance (gm) cells  850  and  860  coupled with mixer/multipliers  855  and  865 , in such a way that the input signal  801  is propagated in a loop, through gm cell  850 , whose output signal  851  is coupled with down-converting mixer  855 , whose output signal  804  is coupled with up-converting mixer  865 . The output signal  861  of up-converting mixer/multiplier  865  is coupled with gm cell  860 , whose output signal  862  (a current) subtracts from input signal  801 . The signal  804  meets filter  840 , which could be coupled in shunt (as shown), or in series, or a combination thereof. Up-converting mixer/multiplier  875  converts signal  803  from baseband or IF to RF frequencies, and injects the reconstructed transmit signal  880  into either gm cells  850  (signal  880   b ), gm cell  860  (signal  880   a ), or directly into the input (signal  880   c ), adding it to the current delivered by the LNA into input  801 . All three mixer/multipliers  855 ,  865 , and  875  are driven by local oscillator  175 . In an alternative embodiment, mixer/multiplier  875  could be driven by local oscillator  135 . The IF or baseband output signal  804  can be taken from the signal input of mixer/multiplier  865 . 
     In embodiments, the reconstruction signal  880  can be injected in one of the gm cells  850  or  860  on the gate in a common-gate design, or it can be injected on the source in a common-source design, or it could be injected directly into the LNA  160  output and CBP mixer  770  input  801 . The reconstructed transmit blocker will need to be aligned in both phase and amplitude with the leaked blocker  510  in order to reduce the impact of the latter. The delay can be controlled with delays in filter  725  or digital processing block  771  prior to the reconstruction DAC  771 , and the amplitude can be controlled by adjusting the gain of the reconstruction DAC  771  or by way of amplifiers in filter  725  or multiplications in digital processing block  771 . The control of both the delay and gain can be achieved by observing the residual leaked transmitted blocker at any point after the CBP mixer  770  and decorrelating the residual error with the baseband transmit signal and delay versions thereof. 
     As discussed above, embodiments of the invention include an adjacent channel leakage canceler (ACL) block to replicate the nonlinear intermodulation (IM) products arising from the transmitted DL signal  310  as the latter goes through various nonlinear components and to subtract the replicated IM products from the receive path signal to improve the signal fidelity of the received UL signal. The operation and inventive aspects of an embodiment of ACL cancelation block  785  can be more clearly understood with the illustration of  FIG. 9 . 
       FIG. 9  shows an embodiment of cancelation block  785 . This embodiment of cancelation block  785  includes two inputs: (i) the digitized corrupted received signal from the ADC  180  and (ii) the digital baseband or low-IF transmit signal (denoted as x[n]) which could come from the baseband transmitter  110 . The cancelation block  785  includes a nonlinear basis function generator  910 . Nonlinear basis function generator  910  receives the digital baseband transmit signal x[n]  905  and outputs a plurality (three are illustrated) of linearly delayed and/or nonlinear basis functions  920   a - c  derived from the digital sequence x[n]. Embodiments of the invention may use a variety of functions, including the multiplication of the digital sequence x[n] (or a derivative thereof) by a complex exponential to account for the difference in frequency between the carrier frequency of the downlink channel and the uplink channel. 
     In one illustrative embodiment, the basis functions could be selected from those from a Volterra series expansion, e.g. a subset of functions with the following pattern: 
               x   ⁡     [   n   ]       ,     
     ⁢     x   ⁡     [     n   -   1     ]       ,     
     ⁢     x   ⁡     [     n   -   2     ]       ,     …   ⁢           ⁢     x   ⁡     [     n   -   M     ]         ,     
     ⁢       x   ⁡     [   n   ]       ^   2     ,     
     ⁢       x   ⁡     [   n   ]       ⁢     x   ⁡     [     n   -   1     ]         ,     
     ⁢       x   ⁡     [   n   ]       ⁢     x   ⁡     [     n   -   2     ]         ,     …   ⁢           ⁢     x   ⁡     [   n   ]       ⁢     x   ⁡     [     n   -   M     ]         ,     
     ⁢       x   ⁡     [     n   -   1     ]       ^   2     ,     
     ⁢       x   ⁡     [     n   -   1     ]       ⁢     x   ⁡     [     n   -   2     ]         ,     …   ⁢           ⁢     x   ⁡     [     n   -   1     ]       ⁢     x   ⁡     [     n   -   M     ]         ,     
     ⁢   …                   x   ⁡     [     n   -   K     ]       ^   2     ,     
     ⁢       x   ⁡     [     n   -   K     ]       ⁢     x   ⁡     [     n   -   K   -   1     ]         ,     …   ⁢           ⁢     x   ⁡     [     n   -   K     ]       ⁢     x   ⁡     [     n   -   M     ]               
for a Volterra series with polynomial order K and memory order M. Besides Volterra series, the nonlinear basis functions could be based on, for instance, Legendre polynomials. A variety of other nonlinear basis functions could be used without deviating from the scope of the invention.
 
     Each of the basis functions  920   a - c  output from the generator  910  are then scaled with a multiplication operation  930   a - c . Amplifiers  930  are drawn to illustrate the multiplication for simplicity, but in practice this could be implemented as a digital multiplication operation. The scaled basis functions are then summed (for example using a digital summation operation illustrated with summation node  940   a ) to generate the replica signal  950  modeling the negative of the intermodulation products  640   a - f . This replica signal  950  can then be added to the output  960  from the ADC  180  to yield an improved received signal  970  which is then provided to the baseband receiver  190 . In a further embodiment, the summation operations  940   a  and  940   b  are combined into a single summation operation where signal  950  would not explicitly exist. 
     The multiplication factor for each of the multiplications  930   a - c  can be adaptively controlled by corresponding control blocks  980   a - c . Each control block  980  can take as input its corresponding basis function  920  and the improved received signal  970  as shown in  FIG. 9 . Then comparing these two signals, it can be determined whether the current scaling is (i) appropriate, (ii) too small and hence needs to be increased, or (iii) too large and hence needs to be decreased. One such means to implement this control block is with least-means-squares (LMS) control or one of its well-known variants. In such a case, the two inputs to control block  980  are effectively correlated and the associated multiplication factor is adjusted in the opposite direction of the correlation until the correlation is zero. 
       FIG. 10  illustrates a conventional radio system  1000  utilizing digital pre-distortion (DPD) in the transmit path. System  1000  observes the output of the power amplifier (PA)  140  with a sensor  1012  and down-converts the sensed PA output with down-converting mixer  1013 , which is controlled by local oscillator  135 . DPD ADC  1014  samples the output of the DPD down-converting mixer  1013  and sends the digitized down-converted PA output to the DPD block  1011  included in baseband transmitter block  1010 . The DPD block  1011  introduces pre-distortion components into the baseband transmitter output so that the overall output of the PA  140  has an improved ACLR. 
     Further embodiments of the invention may combine systems  700   a  and  700   b  with the digital predistortion (DPD) aspects shown in system  1000 .  FIG. 11 a    and embodiment  1100   a  illustrate one such combination. In this example, a transmit signal  1102  from the output of mixer  1013  is sent to filter  725  to construct an approximation of the leakage noise and then sent to CDM  770 . 
     Additionally, the sensed PA output signal, after down-converting and digitizing, is provided to the canceller block  785  input x[n]  905  from the output of DPD ADC  1014 . Such an input would be advantageous in that if the DPD is working well, then the PA output has a very good ACLR with negligible intermodulation products. Thus, the canceller block  785  need only model the distortions introduced by the antenna  155 , LNA  160 , CBP mixer  770 , and ADC  180 . 
     Alternatively, embodiments of the invention may include a DPD system  1100   b  as illustrated in  FIG. 11 b   . As with  FIG. 7 b   , this embodiment allows for digitally processing the transmit blocker  310   a - 310   d  in module  771  and converting it to analog in DAC  772  prior to offering it to CDM/CBP  770 . In this embodiment, both the leakage noise approximation and the ACL cancellation signal are generated from a down-converted and digitized version of the sensed PA output signal taken from the output of DPD ADC  1014 . 
     Although the description has been described with respect to particular embodiments thereof, these particular embodiments are merely illustrative, and not restrictive. Embodiments of the invention may be utilized in conjunction with any type of data encoding and/or modulation scheme known in the art. 
     Embodiments of the invention may be implemented using dedicated hardware and/or software executing on a general purpose computer processor, digital signal processor, stream processing system, application specific integrated circuit (ASIC) or any other type of hardware capable of executing one or more software programs. If software is used to implement any portion of an embodiment of the invention, any suitable programming language can be used to implement the routines of particular embodiments including C, C++, Java, assembly language, etc. Different programming techniques can be employed such as procedural or object oriented. The routines can execute on a single processing device or multiple processors. Although the steps, operations, or computations may be presented in a specific order, this order may be changed in different particular embodiments. In some particular embodiments, multiple steps shown as sequential in this specification can be performed at the same time. 
     Particular embodiments may be implemented in a computer-readable storage medium for use by or in connection with the instruction execution system, apparatus, system, or device. Particular embodiments can be implemented in the form of control logic in software or hardware or a combination of both. The control logic, when executed by one or more processors, may be operable to perform that which is described in particular embodiments. 
     Particular embodiments may be implemented by using a programmed general-purpose digital computer, by using application-specific integrated circuits, programmable logic devices, field programmable gate arrays, etc. Optical, chemical, biological, quantum or nano-engineered systems, components and mechanisms may be used. In general, the functions of particular embodiments can be achieved by any means as is known in the art. Distributed, networked systems, components, and/or circuits can be used. Communication, or transfer, of data may be wired, wireless, or by any other means. 
     It will also be appreciated that one or more of the elements depicted in the drawings/figures can also be implemented in a more separated or integrated manner, or even removed or rendered as inoperable in certain cases, as is useful in accordance with a particular application. It is also within the spirit and scope to implement a program or code that can be stored in a machine-readable medium to permit a computer to perform any of the methods described above. 
     As used in the description herein and throughout the claims that follow, “a”, “an”, and “the” includes plural references unless the context clearly dictates otherwise. Also, as used in the description herein and throughout the claims that follow, the meaning of “in” includes “in” and “on” unless the context clearly dictates otherwise. 
     Thus, while particular embodiments have been described herein, latitudes of modification, various changes, and substitutions are intended in the foregoing disclosures, and it will be appreciated that in some instances some features of particular embodiments will be employed without a corresponding use of other features without departing from the scope and spirit as set forth. Therefore, many modifications may be made to adapt a particular situation or material to the essential scope and spirit.