Abstract:
A converter includes a DC input; a transformer including first and second primary windings, first and second secondary windings, and first and second feedback windings; a first field-effect transistor; a second field-effect transistor; and a drive circuit connected to the first and second field-effect transistors. The drive circuit includes a bias circuit that applies a bias voltage to gates of the first and second field-effect transistors via the first and second feedback windings during start-up of the converter, wherein the bias voltage is reduced to zero or substantially zero after start-up of the converter; and a reset circuit that resets the bias circuit when the converter is turned off. The converter is a self-oscillating push-pull DC-DC converter.

Description:
BACKGROUND OF THE INVENTION 
       [0001]    1. Field of the Invention 
         [0002]    The present invention relates to power conversion, and more specifically, the present invention relates to a circuit and method for driving FETs (field-effect transistors) in a saturating self-oscillating push-pull isolated DC-DC converter. 
         [0003]    2. Description of the Related Art 
         [0004]    Bipolar junction transistors (BJTs) have been used as driving switches in saturating self-oscillating push-pull isolated DC-DC converters. However, because BJTs are current-controlled devices, they generally exhibit larger switching losses than voltage-controlled devices, such as MOSFETs (metal-oxide-semiconductor field-effect transistors). These switching losses become a significant source of power loss in a saturating self-oscillating push-pull isolated DC-DC converter that is operated at relatively high switching frequencies. 
         [0005]    Accordingly, when BJTs are used in a saturating self-oscillating push-pull isolated DC-DC converter, the transformer of the DC-DC converter must be designed with a high number of primary turns and/or a large cross-sectional core area to lower the switching frequency. Furthermore, BJTs typically have gains that vary widely with temperature, which can result in a number of problems, including difficulty during start-up at low operating temperatures. 
       SUMMARY OF THE INVENTION 
       [0006]    The preferred embodiments of the present invention provide a circuit and method that drive switches of a saturating self-oscillating push-pull isolated DC-DC converter with relatively high switching frequencies, which allows the overall size of the transformer to be reduced. Further, the preferred embodiments of the present invention require a relatively small number of turns on the primary winding of the transformer, and switching losses are reduced by the preferred embodiments of the present invention by using FETs, in particular, MOSFETs, that require very little gate-drive current. The preferred embodiments of the present invention also provide stable operation of the DC-DC converter over a wide temperature range, as well as reliable start-up at low temperatures. 
         [0007]    A converter includes a DC input; a transformer including first and second primary windings, first and second secondary windings, and first and second feedback windings; a first field-effect transistor; a second field-effect transistor; and a drive circuit connected to the first and second field-effect transistors. The drive circuit includes a bias circuit that applies a bias voltage to gates of the first and second field-effect transistors via the first and second feedback windings during start-up of the converter, wherein the bias voltage is reduced to zero or substantially zero after start-up of the converter; and a reset circuit that resets the bias circuit when the converter is turned off. The converter is a self-oscillating push-pull DC-DC converter. 
         [0008]    The self-oscillating push-pull DC-DC converter is a Royer oscillator. 
         [0009]    The bias circuit preferably includes a capacitor and a resistor connected in series across the DC input and ground. A first terminal of the capacitor is preferably connected to the DC input. A second terminal of the capacitor is preferably connected to a first terminal of the resistor and a center-tap of the first and second feedback windings. The first terminal of the resistor is preferably connected to the second terminal of the capacitor and the center-tap of the first and second feedback windings. A second terminal of the resistor is preferably connected to ground. The resistor preferably reduces a voltage at the center-tap of the first and second feedback windings to ground or substantially to ground after the capacitor is fully charged. 
         [0010]    The reset circuit preferably includes a resistor connected in parallel with the bias circuit between the DC input and ground. 
         [0011]    The drive circuit preferably further includes a first resistor connected between the gate of the first field-effect transistor and the first feedback winding; and a second resistor connected between the gate of the second field-effect transistor and the second feedback winding. 
         [0012]    The converter preferably further includes a capacitor connected to the first and second secondary windings. 
         [0013]    The converter preferably further includes a rectifier circuit connected to the first and second secondary windings. The rectifier circuit preferably includes first and second diodes. 
         [0014]    The above and other features, elements, characteristics, steps, and advantages of the present invention will become more apparent from the following detailed description of preferred embodiments of the present invention with reference to the attached drawings. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0015]      FIG. 1  is a circuit diagram of a saturating self-oscillating push-pull isolated DC-DC converter according to a preferred embodiment of the present invention. 
           [0016]      FIG. 2  shows oscilloscope traces associated with a transistor in the saturating self-oscillating push-pull isolated DC-DC converter of  FIG. 1 . 
       
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
       [0017]      FIG. 1  shows a saturating self-oscillating push-pull isolated DC-DC converter according to a preferred embodiment of the present invention. The saturating self-oscillating push-pull isolated DC-DC converter of  FIG. 1  is sometimes referred to as a Royer oscillator. As shown in  FIG. 1 , the DC-DC converter includes an input voltage Vin, an output voltage Vout, and a transformer TX 1  with first and second primary windings P 1  and P 2 , first and second secondary windings S 1  and S 2 , and first and second feedback windings F 1  and F 2 . Diodes D 1  and D 2  rectify the voltage of the secondary windings S 1  and S 2 , and an output capacitor C 2  stores energy and smooths the rectified voltage. 
         [0018]    As shown in  FIG. 1 , the DC-DC converter also includes a FET drive circuit with first through fourth resistors R 1  to R 4  and a capacitor C 1  that provides voltages to drive FETs TR 1  and TR 2 . 
         [0019]    During start-up of the DC-DC converter, capacitor C 1  initially begins to charge, which applies a DC bias voltage equal or substantially equal to the input voltage Vin at a center-tap of the feedback winding (i.e., the point between the first and second feedback windings F 1  and F 2 ). A positive DC bias voltage is applied to the gate terminals of the FETs TR 1  and TR 2 , via the feedback windings F 1  and F 2  and the resistors R 3  and R 4 . Due to inherent minor manufacturing differences (e.g., variations within normal manufacturing tolerances) between the FETs TR 1  and TR 2 , one FET TR 1  or TR 2  may have a slightly lower gate threshold voltage V GS(th)  which causes it to switch on first. Here, it is assumed that FET TR 1  has a lower gate threshold voltage V GS(th)  than FET TR 2 , such that FET TR 1  turns on first. 
         [0020]    As FET TR 1  switches on, current flows through the first primary winding P 1  and induces a magnetic field in the transformer TX 1 . This increases the magnetic flux of the transformer TX 1  and results in voltages being induced in the second primary winding P 2 , the first and second secondary windings S 1  and S 2 , and the first and second feedback windings F 1  and F 2 . Accordingly, a negative voltage is induced across the first feedback winding F 1 , and a positive voltage is induced across the second feedback winding F 2 . The positive voltage across the second feedback winding F 2  switches FET TR 1  on, such that FET TR 1  is in saturation mode, and the negative voltage across the first feedback winding F 1  switches FET TR 2  off, such that FET TR 2  is in cut-off mode. During the cut-off mode of a FET, the FET is switched off such that there is no current or substantially no current flowing between the source and drain terminals of the FET. The voltage induced across the second primary winding P 2  does not affect the operation of the DC-DC converter at this time because FET TR 2  is switched off which prevents current from flowing through the second primary winding P 2 . When the capacitor C 1  has fully charged (i.e., after start-up of the circuit), the DC bias voltage at the center-tap of the feedback winding (i.e., the point between the first and second feedback windings F 1  and F 2 ) is reduced to ground or substantially ground through resistor R 1 . 
         [0021]    The drain current I d(TR1)  of FET TR 1  that flows in the first primary winding P 1  increases with the magnetic flux in the core of the transformer TX 1  until the core reaches saturation. When the magnetic flux within the core of the transformer TX 1  reaches its peak value B sat , the inductance of the first primary winding P 1  falls to zero. Because the magnetic flux within the core of the transformer TX 1  stops changing when it reaches its peak value B sat , the magnitude of the voltages across the first and second feedback windings F 1  and F 2  fall to zero or substantially zero. This results in FET TR 1  switching off, and causes the magnetic field induced by the first primary winding P 1  to collapse, causing an opposing change in the magnetic flux of the transformer TX 1 . The switching of the FETs TR 1  and TR 2  then reverses, because a positive voltage is induced across the first feedback winding F 1  and a negative voltage is induced across the second feedback winding F 2 . The positive voltage across the first feedback winding F 1  switches FET TR 2  on, and the negative voltage across the second feedback winding F 2  switches FET TR 1  off. 
         [0022]    Because the gate terminals of the FETs TR 1  and TR 2  are driven by negative voltages when they are switched off, rather than simply applying no voltage or a voltage insufficient to drive the FETs TR 1  and TR 2  into saturation, the gate capacitances of the FETs TR 1  and TR 2  are quickly discharged, which results in the FETs TR 1  and TR 2  switching off quickly.  FIG. 2  shows oscilloscope traces associated with FET TR 1 . As shown in  FIG. 2 , FET TR 1  quickly turns on and off, providing a substantially square wave. 
         [0023]    Current in the first and second primary windings P 1  and P 2  is limited by their DC resistances and by the drain-to-source on-resistance R DS(on)  of the FETs TR 1  and TR 2 . If the output voltage Vout is short-circuited or overloaded, the voltage across the first and second primary windings P 1  and P 2  drops, as the current demand at the load exceeds the current limit provided by the DC resistances of the first and second primary windings P 1  and P 2  and the drain-to-source on-resistance R DS(on)  of the FETs TR 1  and TR 2 . The voltages across the first and second feedback windings F 1  and F 2 , and thus the voltages at the gate terminals of FETs TR 1  and TR 2 , drops according to the lowered voltage across the first and second primary windings P 1  and P 2 . Once the voltage at the gate terminals of the FETs TR 1  and TR 2  drops below both of their respective gate threshold voltages V GS(th) , oscillation in the DC-DC converter ceases with only a minor amount of power being dissipated through resistor R 1 . However, the DC-DC converter only needs to once again go through the above-described start-up sequence to restart. Preferably, this is achieved by simply momentarily disconnecting the input voltage Vin so that capacitor C 1  discharges via resistor R 1 . 
         [0024]    Preferably, the FETs TR 1  and TR 2  have a low drain-to-source on-resistance R DS(on)  to reduce losses and power dissipation. According to preferred embodiments of the present invention, the size of the output capacitor C 2  is selected to significantly reduce or minimize load transients on the output voltage Vout, for example, to prevent the DC-DC converter from undesirably shutting down because of the load transients causing a short-circuit or overload condition. 
         [0025]    The size of capacitor C 1  sets the amount of time for the circuit to start. That is, larger values of the capacitor C 1  increase the time that the positive DC bias voltage is applied to the gate terminals of the FETs TR 1  and TR 2  during start-up of the DC-DC converter. Resistor R 1  discharges capacitor C 1  when the input voltage Vin is disconnected, whereas resistor R 2  allows capacitor C 1  to charge and reduces the DC bias voltage on the gate terminals of FETs TR 1  and TR 2  to zero or substantially zero once capacitor C 1  has charged. Higher values of resistor R 1  and R 2  reduce the quiescent power dissipation of the DC-DC converter. Resistors R 3  and R 4  dampen the voltage applied to drive the gate terminals of FETs TR 1  and TR 2  to reduce gate “ringing” due to the LC circuit defined by the inductances of the first and second feedback windings F 1  and F 2  and the intrinsic gate capacitances of the FETs TR 1  and TR 2 . The values of resistors R 3  and R 4  are preferably large enough to reduce unwanted oscillations, but small enough to avoid excessive clewing of the gate-drive voltage. If the values of the resistors R 3  and R 4  are too high, the FETs TR 1  and TR 2  may be driven in their ‘resistive’ mode, which increases their power dissipation. 
         [0026]    The turns ratios of the primary windings (i.e., the total turns of the first primary winding P 1  and the second primary winding P 2 ) and the feedback windings (i.e., the total turns of the first feedback winding F 1  and the second feedback winding F 2 ) on the transformer TX 1  set the switching frequency and the gate-drive voltage of the FETs TR 1  and TR 2 . The magnitude of the gate-drive voltage V gs  is determined by the equation: 
         [0000]    
       
         
           
             
               
                  
                 
                   V 
                   gs 
                 
                  
               
               = 
               
                 
                   
                     V 
                     p 
                   
                   × 
                   
                     N 
                     f 
                   
                 
                 
                   N 
                   p 
                 
               
             
             , 
           
         
       
     
         [0000]    and the switching frequency f can be calculated by the equation: 
         [0000]    
       
         
           
             
               f 
               = 
               
                 
                   V 
                   p 
                 
                 
                   4 
                   × 
                   B 
                   × 
                   
                     N 
                     p 
                   
                   × 
                   
                     A 
                     e 
                   
                 
               
             
             , 
           
         
       
     
         [0000]    where V p  is the voltage at the primary winding, N f  is the number of feedback turns, N p  is the number of primary turns, B is the peak flux density of the transformer core, and A e  is the effective cross-sectional area of the transformer core. The above equation for the switching frequency f is determined with respect to one-half of the primary winding, i.e., only one of the first primary winding P 1  and the second primary winding P 2 . However, the first primary winding P 1  preferably has the same number of turns as the second primary winding P 2 . Further, the first secondary winding S 1  preferably has the same number of turns as the second secondary winding S 2 , and the first feedback winding F 1  preferably has the same number of turns as the second feedback winding F 2 . 
         [0027]    Accordingly, the DC-DC converter is able to operate at relatively high switching frequencies, although the range of switching frequencies is limited by various factors. For example, the range switching frequencies for the DC-DC converter is also limited by core losses, the values of resistors R 3  and R 4 , the gate capacitances of the FETs TR 1  and TR 2 , and the like. 
         [0028]    According to the preferred embodiments of the present invention, the drain-to-source on-resistance R DS(on)  of each of the FETs TR 1  and TR 2  has a positive thermal coefficient to provide a self-balancing effect in the push-pull arrangement of the DC-DC converter, stable operation of the DC-DC converter over a wide temperature range, and reliable start-up of the DC-DC converter even at low temperatures. 
         [0029]    The preferred embodiments of the present invention can be applied to both N-channel FETs and P-channel FETs. If P-channel FETs are used, the supply rails of the input voltage Vin are reversed. 
         [0030]    According to the preferred embodiments of the present invention, Zener diodes can be placed in parallel with gate and source terminals of the FETs TR 1  and TR 2  to clamp the maximum gate-drive voltage Vgs. 
         [0031]    It should be understood that the foregoing description is only illustrative of the present invention. Various alternatives and modifications can be devised by those skilled in the art without departing from the present invention. Accordingly, the present invention is intended to embrace all such alternatives, modifications, and variances that fall within the scope of the appended claims.