Abstract:
A power supply control device, a power supply control circuit, a power supply control method, and an electronic apparatus wherein a determination is made of whether or not a switching element therein is in an ON state. When the switching element is in the ON state, a voltage generated across the switching element is detected in accordance with a current flowing through the switching element. In accordance with the detected voltage, the switching element is controlled and thereby an output voltage is controlled.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention generally relates to power supply control devices, power supply circuits, power supply control methods, and electronic apparatuses. More particularly, the present invention relates to a power supply control device, a power supply circuit, a power supply control method, and an electronic apparatus, in which current to be supplied to a load is controlled by a transistor disposed between a power source and the load. 
     2. Description of the Related Art 
     In recent years, portable electronic apparatuses have been widespread. A portable electronic apparatus operates with a battery unit as its power source. Accordingly, the operating time of the apparatus by the battery unit is an important factor in measuring the performance of the apparatus. Such a portable electronic apparatus does not drive an internal circuit with the voltage of the battery unit. Therefore, a power supply circuit that converts the voltage of the battery unit into a voltage suitable for the internal circuit is contained in the apparatus. 
     It is preferable that the power consumption of the apparatus be reduced so as to prolong the operation time, while high efficiency is maintained in the power supply circuit. However, a high-efficient power supply circuit consumes a large amount of power. 
     Conventionally, synchronous DC-DC converters are used to improve the efficiency of a power supply circuit. As the performance of the CPUs contained in electronic apparatuses has been increased, the power consumption has been becoming larger. To restrict such an increase in power consumption, the voltage used in an electronic apparatus has been decreasing. As a result, the output of a DC-DC converter has become low-voltage, large-current output. 
     In view of this, it is essential to protect a DC-DC converter when short-circuiting or overloading occurs. Example methods of protecting a DC-DC converter from short-circuiting and overloading includes: a constant current control method in which the maximum value of the output current of the DC-DC converter is restricted through monitoring the output current; and an excess current preventing circuit method in which, the instant an excess current is detected, the DC-DC converter is stopped. 
     Generally, a current sense resistor is disposed in the output circuit of the DC-DC converter so as to monitor the voltage caused by the current flowing through such a current sense resistor. In a portable electronic apparatus, a battery unit is used as a power source for the internal devices. As a battery keeps discharging, the voltage of the battery normally drops. Therefore, DC-DC converters are employed to maintain the voltage used in an electronic apparatus at a constant value. 
     In a notebook-type computer, for instance, various devices such as a semiconductor device, a storage device, and a display device are mounted. These devices operate at different voltages from each other. More specifically, devices such as a HDD, CD-ROM, and DVD operate at 5.0 V, while a memory and a semiconductor device for controlling peripheral circuits operate at 3.3 V, for instance. As for the CPU, the operating voltage is 0.9 to 2.0 V. 
     Meanwhile, power is supplied to such a notebook-type computer from an external power source such as an AC adapter, or from a battery unit contained in the apparatus. In this case, a DC-DC converter is employed to produce various voltages required by the devices in the apparatus. 
     For such a DC-DC converter, a switching regulator type is widely used for its high efficiency. In a DC-DC converter of the switching regulator type, a transistor is disposed between a power source and a load, and this transistor is controlled so as to control the output voltage. To detect the current to be supplied to the load, a current sense resistor is connected in series between the transistor and the load. As the current to be supplied to the load flows through the current sense resistor, a voltage corresponding to the current is generated across the current sense resistor. By detecting this voltage generated across the current sense resistor, the current to be supplied to the load can be detected. Using the detected current, a control operation such as an excess current preventing operation is performed. 
     FIG. 1 is a block diagram of a power system of an electronic apparatus. 
     In a portable electronic apparatus  1 , such as a notebook computer, a commercial AC power supply  3  is converted into a DC power supply by an AC adapter  2 , and the DC power supply is generally used as the driving power. The conversion is externally carried out. The electronic apparatus  1  comprises a battery unit  5 , a charger  6 , diodes D 11  and D 12 , and DC-DC converters  7 - 1  to  7 - 3 . With this structure, the DC power from the AC adapter  2  is supplied to internal units  4 - 1  to  4 - 3 . 
     The battery unit  5  is used as the driving power when the electronic apparatus  1  is being carried. The charger  6  recharges the battery unit  5  with external power from the AC adapter  2 . 
     The diode D 11  prevents power supply from the battery unit  5  to the AC adapter  2 . The diode D 12  prevents direct voltage application from the AC adapter  2  to the battery unit while the AC adapter  2  is connected to the electronic apparatus  1 . 
     The DC-DC converter  7 - 1  converts DC voltage from the AC adapter  2  or the battery unit  5  into DC voltage demanded by the internal unit  4 - 1 , and then supplies the converted DC voltage to the internal unit  4 - 1 . The DC-DC converter  7 - 2  converts DC voltage from the AC adapter  2  or the battery unit  5  into DC voltage demanded by the internal unit  4 - 2 , and then supplies the converted DC voltage to the internal unit  4 - 2 . The DC-DC converter  7 - 3  converts DC voltage from the AC adapter  2  or the battery unit  5  into DC voltage demanded by the internal unit  4 - 3 , and then supplies the converted DC voltage to the internal unit  4 - 3 . 
     FIG. 2 is a block diagram of conventional DC-DC converters. 
     The DC-DC converters  7 - 1  to  7 - 3  each comprise a power supply control IC  10 , a main switching transistor Tr 1 , a synchronous rectifying transistor Tr 2 , diodes D 1  and D 2 , a choke coil L 1 , a smoothing capacitor C 1 , a back-flow preventing capacitor C 2 , and a current sense resistor R 1 . 
     An input voltage Vin is supplied to an input terminal Tin. The input terminal Tin is connected to the power supply terminal Tvin of the power supply control IC  10  and the drain of the main switching transistor Tr 1 . 
     The main switching transistor Tr 1  is constituted by an n-channel MOSFET (Metal Oxide Semiconductor Field Effect Transistor). The drain of the main switching transistor Tr 1  is connected to the input terminal Tin, while the source is connected to an output terminal Tout via the choke coil L 1  and the current sense resistor R 1 . The gate of the main switching transistor Tr 1  is connected to a terminal Tdh of the power supply control IC  10 . The main switching transistor Tr 1  is switched on and off, depending on a pulse supplied from the terminal Tdh of the power supply control IC  10 . 
     The output current of the main switching transistor Tr 1  is supplied to the choke coil L 1 . The choke coil L 1 , the synchronous commutating transistor Tr 2 , and the diode D 2  constitute a rectifier circuit that rectifies the pulse-type output current from the main switching transistor Tr 1 . 
     The anode of the diode D 1  is grounded, and the cathode of the diode D 1  is connected to the choke coil L 1 . The diode D 1  is a flywheel diode that supplies forward current to the choke coil L 1  when the main switching transistor Tr 1  is switched off, and is provided with reverse voltage and switched off when the main switching transistor Tr 1  is on. 
     The synchronous rectifying transistor Tr 2  is constituted by an n-channel MOSFET (Metal Oxide Semiconductor Field Effect Transistor). The drain of the synchronous rectifying transistor Tr 2  is connected to the source of the main switching transistor Tr 1 , while the source thereof is grounded. The gate of the synchronous rectifying transistor Tr 2  is connected to a terminal Td 1  of the power supply control IC  10 . The synchronous rectifying transistor Tr 2  is switched off by the power supply control IC  10  when the main switching transistor Tr 1  is switched on, and is switched on by the power supply control IC  10  when the main switching transistor Tr 1  is switched off. The synchronous rectifying transistor Tr 2  is switched on when the forward current flows through the diode D 1 , so as to reduce a voltage decrease due to the forward voltage of the diode D 1 . 
     The current rectified by the choke coil L 1 , the diode D 1 , and the synchronous transistor Tr 2  is supplied to the output terminal Tout via the resistance R 1  of the current sense resistor R 1 . The connection point between the choke coil L 1  and the current sense resistor R 1  is connected to a terminal Tcs of the power supply control IC  10 . 
     The output terminal Tout is connected to the ground via the smoothing capacitor C 1 , as well as to a terminal Tfb of the power supply control IC  10 . The smoothing capacitor C 1  smoothens current supplies from the choke coil L 1  via the current sense resistor R 1 . 
     Furthermore, a driving power terminal Tvg is connected to a terminal Tvb of the power supply control IC  10 . A gate driving voltage VG is applied to the driving power terminal Tvg. A terminal Tcb of the power supply control IC  10  is connected to the connection point between the diode D 2  and the back-flow preventing capacitor C 2 . The anode of the diode D 2  is connected to the driving power terminal Tvg, while the cathode thereof is connected to the back-flow preventing capacitor C 2 . The diode D 2  prevents the current from flowing back toward the terminal Tvg when the voltage of a terminal Tcb is increased by the back-flow preventing capacitor C 2 . One end of the back-flow preventing capacitor C 2  is connected to the connection point between the cathode terminal of the diode D 2  and the terminal Tcb, while the other end thereof is connected to the connection point of the source of the main switching transistor Tr 1 , the drain of the synchronous rectifying transistor Tr 2 , the cathode of the diode D 1 , and one end of the choke coil L 1 . 
     Referring now to FIG. 3, the power supply control IC  10  will be described in greater detail. 
     The power supply control IC  10  comprises a differential amplifier  11  (AMP 1 ), error amplifiers  12  and  13  (ERA 2  and ERA 1 ), a saw-tooth oscillator  14 , a PWM, (Pulse Width Modulation) comparator  15 , drive amplifiers  16  and  17  (DRV 1  and DRV 2 ), reference voltage source  18  and  19 , and resistors R 2  and R 3 . 
     The non-inverting input terminal of the differential amplifier  11  is connected to the terminal Tcs, while the inverting input terminal thereof is connected to the terminal Tfb. Accordingly, the differential amplifier  11  outputs a differential signal in accordance with the potential difference between the terminal Tcs and the terminal Tfb. The terminal Tcs and the terminal Tfb are connected to both ends of the current sense resistor R 1  outside the power supply control IC  10 . Accordingly, the differential signal outputted from the differential amplifier  11  depends on the current flowing through the current sense resistor R 1 . 
     The differential signal outputted from the differential amplifier  11  is supplied to the inverting input terminal of the error amplifier  12 . A reference voltage e 2  generated from the reference voltage source  18  is applied to the non-inverting input terminal of the error amplifier  12 . The error amplifier outputs a signal that varies depending on the difference between the differential signal outputted from the differential amplifier  11  and the reference voltage e 2 . Accordingly, the output of the error amplifier  12  is small when the current flowing through the current sense resistor R 1  is great. On the other hand, the output of the error amplifier  12  is great when the current flowing through the current sense resistor R 1  is small. The output signal of the error amplifier  12  is supplied to a non-inverting input terminal of the PWM comparator  15 . 
     A divided voltage between the resistors R 2  and R 3  is supplied to the inverting input terminal of the error amplifier  13 , while a reference voltage e 1  of the reference voltage source  19  is supplied to the non-inverting input terminal. The resistors R 2  and R 3  are connected in series between the terminal Tfb and the terminal Tgnd, so as to divide the voltage applied to the terminal Tfb. The terminal Tfb is connected to the output terminal Tout outside the power supply control IC  10 . The resistors R 2  and R 3  divide the output voltage Vout, and supply it to the error amplifier  13 . 
     The error amplifier  13  outputs a value that is obtained by subtracting the divided voltage between the resistors R 2  and R 3  from the reference voltage e 1  generated from the reference voltage source  19 . When the output voltage Vout of the output terminal Tout is small, the output of the error amplifier  13  is great. On the other hand, when the output voltage Vout of the output terminal Tout is great, the output of the error amplifier  13  is small. The output of the error amplifier  13  is supplied to a non-inverting input terminal of the PWM comparator  15 . 
     A saw-tooth signal is supplied from the saw-tooth oscillator  14  to an inverting input terminal of the PWM comparator  15 . The PWM comparator  15  compares the saw-tooth signal with the signals outputted from the error amplifiers  12  and  13 , so as to output a pulse in accordance with the comparison result. The PWM comparator  15  compares one of the outputs of the error amplifier  12  and  13 , whichever is smaller, with the saw-tooth wave generated from the saw-tooth oscillator  14 . The PWM comparator then outputs a pulse that is high when the smaller output between the error amplifiers  12  and  13  is smaller than the saw-tooth wave, and low when the smaller output between the error amplifiers  12  and  13  is greater than the saw-tooth wave. 
     FIG. 4 is an operation waveform chart of a conventional power supply control IC when performing voltage control. FIG. 5 is an operation waveform chart of another conventional power supply control IC when performing current control. In FIGS. 4 and 5, (A) shows the output waveforms of the error amplifiers  12  and  13 , and the saw-tooth oscillator  14 , and (B) shows the output waveform of the PWM comparator  15 . 
     When the output current Iout is relatively small and the output voltage Vout is high, the output ERA 1  of the error amplifier  13  and the saw-tooth wave are compared, as shown in FIG.  4 A. The pulse outputted from the PWM comparator  15  varies in duty ratio of the high level to the low level, depending on the output ERA 1  of the error amplifier  13 , as shown in FIG.  4 B. When the output voltage Vout is high, the pulse outputted from the PWM comparator  15  has a smaller pulse width at the high level and a greater pulse width at the low level. When the output voltage Vout is low, the pulse outputted from the PWM comparator  15  has a greater pulse width at the high level and has a smaller pulse width at the low level. 
     When the output voltage Vout is relatively low and the output current Iout is great, the output ERA 2  of the error amplifier  12  and the saw-tooth wave are compared, as shown in FIG.  5 A. The pulse outputted from the PWM comparator  15  varies in duty ratio of the high level to the low level, depending on the output ERA 2  of the error amplifier  12 , as shown in FIG.  5 B. When the output current Iout is great, the pulse outputted from the PWM comparator  15  has a smaller pulse width at the high level and has a greater pulse width at the low level. When the output current Iout is small, the pulse outputted from the PWM comparator  15  has a greater pulse width at the high level and has a smaller pulse width at the low level. 
     The PWM comparator  15  supplies its output pulse to the drive amplifier  16 . The drive amplifier  16  is connected to the terminal Tcb and the terminal Tv 1 , and is driven in accordance with the potential difference between the terminal Tcb and the terminal Tv 1 . The drive amplifier  16  generates a driving signal from the output pulse of the PWM comparator  15 . The driving signal is used for driving the main switching transistor Tr 1 . The output of the drive amplifier  16  is outputted through the terminal Tdh. The terminal Tdh of the power supply control IC  10  is connected to the gate of the main switching transistor Tr 1 , so that a pulse that depends on the output pulse of the PWM comparator  15  can be supplied from the terminal Tdh to the gate of the main switching transistor Tr 1 . 
     The main switching transistor Tr 1  is switched on when the pulse supplied from the terminal Tdh is high, and is switched off when the pulse supplied from the terminal Tdh is low. 
     Aside from the output pulse, the PWM comparator  15  also outputs an inverted output pulse that is obtained by inverting the output pulse. The inverted output pulse is supplied to the drive amplifier  17 . The drive amplifier  17  is connected to the terminal Tvb, and is driven by the driving voltage VG supplied to the terminal Tvb. From the inverted output pulse, the drive amplifier  17  generates a driving signal for driving the synchronous rectifying transistor Tr 2 . The output of the drive amplifier  17  is outputted through the terminal Td 1  of the power supply control IC  10 . The terminal Td 1  of the power supply control IC  10  is connected to the gate of the synchronous rectifying transistor Tr 2 , so that a pulse that depends on the inverted output pulse can be supplied to the gate of the synchronous rectifying transistor Tr 2 . The synchronous rectifying transistor Tr 2  is switched on when the pulse supplied from the terminal Td 1  is high, and is switched off when the pulse supplied from the terminal Td 1  is low. 
     The pulse supplied to the gate of the synchronous rectifying transistor Tr 2  is equal to a pulse obtained by inverting the pulse supplied to the gate of the main switching transistor Tr 1 . Accordingly, when the main switching transistor Tr 1  is in the ON state, the synchronous rectifying transistor Tr 2  is in the OFF state. On the other hand, when the main switching transistor Tr 1  is in the OFF state, the synchronous rectifying transistor Tr 2  is in the ON state. 
     In the above power supply control IC  10 , the following relationship is satisfied: 
     
       
           V out=[ T on/( T on+ T off)]× V in=( T on/ T   0 )× V in 
       
     
     wherein Vout is the output voltage, Ton is the duration of the main switching transistor in the ON state, Toff is the duration of the main switching transistor Tr 1  in the OFF state, and (Ton+Toff)=T 0 . 
     The following relationship is also satisfied: 
       I in=( T on/ T   0 )× I out 
     wherein Iin is the average input current, and Iout is the output current. 
     In accordance with the above expressions, the output voltage Vout and the output current Iout can be constantly controlled by controlling the duty ratio in the main switching transistor Tr 1  with the power supply control IC  10 . 
     As described so far, the power supply control IC  10  having the above structure measures the output voltage Vout and the output current Iout. In accordance with the comparison between the output voltage Vout and the output current Iout, the main switching transistor Tr 1  and the synchronous rectifying transistor Tr 2  are controlled. However, the control operation is performed in accordance with the output voltage Vout, and power supply control ICs that perform a current control operation in an excess current state. 
     FIG. 6 is a block diagram of another conventional power supply control IC. In this figure, the same components as in FIG. 3 are denoted by the same reference numerals, and an explanation for those components is omitted. 
     A power supply control IC  21  comprises a comparator  22 , a reference voltage source  23 , a flip-flop  24 , an OR gate  25 , and an AND gate  26 , in place of the differential amplifier  11 , the error amplifier  12 , and the reference voltage source  18  of FIG.  3 . 
     The non-inverting input terminal of the comparator  22  is connected to the terminal Tcs, while the inverting input terminal thereof is connected to the terminal Tfb via the reference voltage source  23 . The reference voltage source  23  generates a reference voltage e 11 . When the voltage at either end of the current sense resistor R 1  is higher than the reference voltage e 11 , the comparator  22  outputs a high-level signal. On the other hand, when the voltage at either end of the current sense resistor R 1  is lower than the reference voltage e 11 , the comparator  22  outputs a low-level signal. The flip-flop  24  is an R-S flip-flop. The output of the comparator  22  is supplied to the set terminal S of the flip-flop  24 , and the saw-tooth wave outputted from the saw-tooth oscillator  14  is supplied to the reset terminal R of the flip-flop  24 . 
     When the output of the comparator  22  becomes high, the flip-flop  24  is set. When the saw-tooth wave outputted from the saw-tooth oscillator  14  reaches a predetermined level, the flip-flop  24  is reset. Once the flip-flop  24  is set, it keeps outputting a high-level signal until it is reset. The flip-flop  24  outputs a non-inverted output Q and an inverted output *Q. The non-inverted output Q is supplied to the OR gate  25 , while the inverted output *Q is supplied to the AND gate  26 . 
     The non-inverted output Q of the flip-flop  24  and the inverted output of the PWM comparator  15  are supplied to the OR gate  25 . The OR gate  25  performs an OR logic operation on the non-inverted output Q of the flip-flop  24  and the inverted output of the PWM comparator  15 . The output signal of the OR gate  25  is supplied to the drive amplifier  17 . 
     Meanwhile, the inverted output *Q of the flip-flop  24  and the non-inverted output of the PWM comparator  15  are supplied to the AND gate  26 . The AND gate  26  performs an AND logic operation on the inverted output *Q of the flip-flop  24  and the non-inverting output of the PWM comparator  15 . The output signal of the AND gate  26  is supplied to the drive amplifier  16 . 
     When the voltage generated at either end of the current sense resistor R 1  is higher than the reference voltage e 11 , the flip-flop  24  is set. When the flip-flop  24  is set, the non-inverted output of the flip-flop  24  becomes high, and the non-inverted output thereof becomes low. 
     When the non-inverted output of the flip-flop  24  becomes high, the output of the OR gate  25  also becomes high, regardless of whether the inverted output of the PWM comparator  15  is high or low. Accordingly, the output of the terminal Td 1  remains high. While the output of the terminal Td 1  is high, the synchronous rectifying transistor Tr 2  remains in the ON state. 
     When the inverted output of the flip-flop  24  becomes low, the output of the AND gate  26  also becomes low, regardless of whether the non-inverted output of the PWM comparator  15  is high or low. Accordingly, the output of the terminal Tdh remains low. While the output of the terminal Tdh is low, the main switching transistor Tr 1  remains in the OFF state. In this manner, the output current Iout is restricted so as to eliminate excess current. 
     When the output of the saw-tooth oscillator  14  reaches a predetermined level, the flip-flop  24  is reset. If an excess current has been eliminated at this point, the flip-flop  24  remains in the reset state, and the regular voltage control is performed. On the other hand, if an excess current has not been eliminated at this point, the flip-flop  24  is set again, and the main switching transistor Tr 1  remains in the OFF state, thereby continuing the excess current eliminating operation. 
     In this structure, the main switching transistor Tr 1  is switched off when excess current is detected, and the voltage control operation is performed when the excess current is eliminated. However, once the main switching transistor Tr 1  is switched off when excess current is detected, it may remain in the OFF state until power supply is resumed, even if the excess current is eliminated. 
     FIG. 7 is a block diagram of yet another conventional power supply control IC. In this figure, the same components as in FIG. 6 are denoted by the same reference numerals. 
     A power supply control IC  31  is characterized by the flip-flop  24  that is not reset by the saw-tooth wave generated from the saw-tooth oscillator  14 . 
     In any of the conventional power supply control ICs described so far, the current sense resistor is connected in series between the main switching transistor and a load. As a result, there is a problem of power loss caused by the current sense resistor. Such power loss becomes larger as the output current becomes greater. 
     To reduce power loss caused by the current sense resistor, the current sense resistor should be made smaller. However, even if the resistance value of the current sense resistor is very small, the power loss cannot be completely avoided. Also, a resistor having a small resistance value is costly. Furthermore, the current sense resistor is normally formed by a discrete component, which takes up considerable space. 
     SUMMARY OF THE INVENTION 
     A general object of the present invention is to provide power supply control devices in which the above disadvantages are eliminated. 
     A more specific object of the present invention is to provide a power supply control device, a power supply circuit, a power supply control method, and an electronic apparatus that are not costly, do not take up too much space, and can reduce power loss. 
     The above objects of the present invention are achieved by a power supply control device which determines whether or not a switching element is in an ON state, detects a voltage generated across the switching element in accordance with a current flowing through the switching element when the switching element is in the ON state, and controls the switching element in accordance with the detected voltage, thereby controlling an output voltage. 
     The above objects of the present invention are also achieved by a power supply circuit, a power supply control method, or an electronic apparatus, which employs the above power supply control device. 
     In this power supply control device, whether or not the switching element is in the ON state is determined in accordance with a control signal for controlling the switching element. Alternatively, whether or not the switching element is in the ON state is determined in accordance with a result of a comparison between the gate-source voltage of the switching element and a reference voltage. In yet another embodiment, the potential difference between the output voltage and a first reference voltage is detected, and whether or not the switching element is in the ON state is determined in accordance with a result of a comparison between the detected potential difference and a second reference voltage. The second reference voltage can be set based on an externally supplied voltage. 
     In accordance with the present invention, an output current can be detected using the ON-resistance of the switching element, thereby eliminating the need for a current sense resistor and power loss due to such a current sense resistor. Thus, the power consumption and costs can be reduced. 
     Furthermore, the voltage for detecting an ON state is set at a suitable value by varying the external voltage, so as to determine whether or not the switching element is certainly in the ON state. 
    
    
     The above and other objects and features of the present invention will become more apparent from the following description taken in conjunction with the accompanying drawings. 
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram of a conventional power system of an electronic apparatus; 
     FIG. 2 is a block diagram of a conventional DC-DC converter; 
     FIG. 3 is a block diagram of a power supply control IC of the conventional DC-DC converter of FIG. 2; 
     FIGS. 4A and 4B show example operation waveforms of the conventional power supply control IC in a voltage control operation; 
     FIGS. 5A and 5B show example operation waveforms of the conventional power supply control IC in a current control operation; 
     FIG. 6 is a block diagram of another conventional power supply IC; 
     FIG. 7 is a block diagram of yet another conventional power supply IC; 
     FIG. 8 is a block diagram of a DC-DC converter as one embodiment of the present invention; 
     FIG. 9 is a block diagram of a power supply control IC of the DC-DC converter of FIG. 8; 
     FIGS. 10A to  10 C illustrate an operation of the power supply control IC of FIG. 9; 
     FIG. 11 is a block diagram of a first modification of the power supply control IC of the present invention; 
     FIG. 12 is a block diagram of a second modification of the power supply control IC of the present invention; 
     FIGS. 13A to  13 C illustrate an operation of the second modification of the power supply control IC of the present invention; 
     FIG. 14 is a block diagram of a third modification of the power supply control IC of the present invention; 
     FIG. 15 is a block diagram of a fourth modification of the power supply control IC of the present invention; and 
     FIG. 16 is a block diagram of a fifth modification of the power supply control IC of the present invention. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The following is a description of embodiments of the present invention, with reference to the accompanying drawings. 
     FIG. 8 is a block diagram of a DC-DC converter as one embodiment of the present invention. In this figure, the same components as in FIG. 2 are denoted by the same reference numerals, and explanations for those components are omitted. 
     A DC-DC converter  100  of this embodiment is used as any of the DC-DC converters  7 - 1  to  7 - 3  contained in the electronic apparatus shown in FIG.  1 . In this DC-DC converter  100 , the current sense resistor R 1  is substituted for by the ON-resistance of the main switching transistor Tr 1 . The DC-DC converter  100  of this embodiment comprises a resistor R 4 , instead of the current sense resistor R 1 , and also comprises a power supply control IC  110  that has a different structure from the power supply control IC  10  shown in FIG.  3 . 
     The power supply control IC  110  comprises a terminal Ti 1 , instead of the terminal Tcs of the power supply control IC  10  shown in FIG.  2 . The resistor R 4  is disposed between the terminal Ti 1  and the terminal Tvin of the power supply control IC  110 . 
     FIG. 9 is a block diagram of the power supply control IC  110  of this embodiment of the present invention. In this figure, the same components as in FIG. 6 are denoted by the same reference numerals, and explanations for those components are omitted. 
     The power supply control IC  110  comprises comparators  121  and  122  in place of the comparator  22  and the reference voltage source  23 , an AND gate  123 , a reference voltage source  124 , and a constant current source  125 . 
     The non-inverting input terminal of the comparator  121  is connected to the terminal Tdh, while the inverting input terminal thereof is connected to the terminal Tv 1  via the reference voltage source  124 . The comparator  121  outputs a signal that is high when the gate-source voltage of the main switching transistor Tr 1  is higher than a reference voltage e 21  of the reference voltage source  124 . The signal outputted from the comparator  121  is low when the gate-source voltage is lower than the reference voltage e 21  of the reference voltage source  124 . In accordance with the comparison result of the comparator  121 , whether or not the main switching transistor Tr 1  is in the ON state can be determined. 
     The non-inverting input terminal of the comparator  122  is connected to the terminal Ti 1 , while the inverting input terminal thereof is connected to the terminal Tv 1 . The constant current source  125  is disposed between the terminal Ti 1  and the ground. The constant current source  25  draws current through the resistor R 4  so as to generate a reference voltage e 31  across the resistor R 4 . The comparator  122  outputs a signal that is high when the drain-source voltage of the main switching transistor Tr 1  is higher than the reference voltage e 31 . The signal outputted from the comparator  122  is low when the drain-source voltage of the main switching transistor Tr 1  is lower than the reference voltage e 31 . In accordance with the comparison result of the comparator  122 , the output current can be judged. 
     The outputs of the comparators  121  and  122  are supplied to the AND gate  123 . The AND gate  123  performs an AND logic operation on the outputs of the comparators  121  and  122 . 
     The output of the AND gate  123  is supplied to the set terminal of the flip-flop  24 . The flip-flop  24  is set when the output of the comparator  121  is high, i.e., the main switching transistor Tr 1  is in the ON state, and the output of the comparator  122  is high, i.e., the drain-source current of the main switching transistor Tr 1  is greater than a predetermined value. 
     FIGS. 10A to  10 C illustrate an operation of the power supply control IC  110  of this embodiment. More specifically, FIG. 10A shows the output of the error amplifier  13  and the saw-tooth wave outputted from the saw-tooth oscillator  14 . FIG. 10B shows the gate-source voltage Vgs of the main switching transistor Tr 1 . FIG. 10C shows the drain-source resistance Rds of the main switching transistor Tr 1 . 
     When the saw-tooth wave outputted from the saw-tooth oscillator  14  becomes smaller than the output of the error amplifier at time t 1 , the output of the PWM comparator  15  becomes high. As the output of the PWM comparator  15  becomes high, the output of the drive amplifier  16  becomes high. The output of the drive amplifier  16  is supplied to the gate of the main switching transistor Tr 1  via the terminal Tdh. 
     As the gate of the main switching transistor Tr 1  becomes high, the gate-source voltage Vgs of the main switching transistor Tr 1  rises as shown in FIG.  10 B. As the gate-source voltage Vgs rises, the main switching transistor Tr 1  is switched on, and the drain-source resistance Rds drops as shown in FIG.  10 C. 
     When the gate-source voltage Vgs reaches an ON-voltage Von at time t 2 , the drain-source resistance Rds of the main switching transistor Tr 1  reaches the minimum value, as shown in FIG.  10 C. Accordingly, the ON-state of the main switching transistor Tr 1  can be accurately detected by setting the reference voltage e 21  of the reference voltage source  124  at a value greater than the ON-voltage Von. Thus, a current value can be invariably detected with a constant value. 
     In this embodiment, when the main switching transistor Tr 1  is certainly in the ON state, excess current can be detected by detecting the current flowing through the main switching transistor Tr 1 , using the ON-resistance of the main switching transistor Tr 1 . Accordingly, the current sense resistor can be omitted, and excessive power consumption can be avoided. 
     Although the reference voltage source  124  is used to detect the ON-state of the main switching transistor Tr 1  in this embodiment, a divided voltage of the drain-source voltage of the main switching transistor Tr 1  may be used, instead of the reference voltage source  124 . 
     FIG. 11 is a block diagram of a first modification of the power supply control IC  110  of the present invention. In this figure, the same components as in FIG. 9 are denoted by the same reference numerals, and explanations for those components are omitted in the following description. 
     A power supply control IC  130  of this modification comprises resistors R 5  and R 6  in place of the reference voltage source  124 . The resistors R 5  and R 6  are connected in series between the terminal Tcb and the terminal Tv 1 , and the connection point of the resistors R 5  and R 6  is connected to the inverting input terminal of the comparator  121 . 
     In this modification, the voltage of the terminal Tcb, which is connected to the external power terminal Tvg via the diode D 2 , is divided by the resistors R 5  and R 6 , and the divided voltage may be adjusted by varying the voltage VG applied to the external power terminal Tvg. Thus, the voltage detected in the comparator  122  can be varied. 
     FIG. 12 is a block diagram of a second modification of the power supply control IC of the present invention. In this figure, the same components as in FIG. 9 are denoted by the same reference numerals, and explanations for those components are omitted in the following description. 
     In a power supply control IC  140  of this modification, the non-inverting input terminal of the comparator  121  is connected to the output of the error amplifier  13 . 
     In this modification, when the output of the error amplifier  13  is greater than the reference voltage e 31  of the reference voltage source  124 , excess current can be detected. Here, the main switching transistor Tr 1  is in the ON-state when the flip-flop  24  is not in the reset state. 
     FIG. 13 shows an operation of the second modification of the power supply control IC of the present invention. More specifically, FIG. 13A shows the output of the error amplifier  13  and the saw-tooth wave outputted from the saw-tooth oscillator  14 . FIG. 13B shows the gate-source voltage Vgs of the main switching transistor Tr 1 . FIG. 13C shows the drain-source resistance Rds of the main switching transistor Tr 1 . 
     If the output of the error amplifier  13  is too low as shown in FIG. 13A, the period of time during which the gate-source voltage vgs of the main switching transistor Tr 1  is an ON-voltage is very short as shown in FIG.  13 B. As a result, the time required for the drain-source resistor Rds of the main switching transistor Tr 1  to be the smallest value cannot be maintained, and a drop of the voltage flowing through the main switching transistor Tr 1  cannot be accurately measured. 
     As can be seen from the above fact, when the output of the error amplifier  13  is greater than the reference voltage e 31  of the reference voltage source  124 , an ON-time can be sufficiently maintained for the main switching transistor Tr 1 . Accordingly, a decrease of the voltage flowing through the main switching transistor Tr 1  can be accurately measured. 
     In the foregoing embodiments, each of the power supply control ICs  110 ,  130 , and  140  performs an operation current control operation. However, these power supply control ICs  110 ,  130 , and  140  may perform an excess current control operation. 
     FIG. 14 is a block diagram of a third modification of the power supply control IC of the present invention. In this figure, the same components as in FIG. 9 are denoted by the same reference numerals, and explanations for those components are omitted in the following description. 
     A power supply control IC  150  of this modification differs from the power supply control IC  110  of FIG. 9 in that the flip-flop  24  is not reset. 
     FIG. 15 is a block diagram of a fourth embodiment of the power supply control IC of the present invention. In this figure, the same components as in FIG. 11 are denoted by the same reference numerals, and explanations for those components are omitted in the following description. 
     A power supply control IC  160  of this modification differs from the power supply control IC  130  of FIG. 11 in that the flip-flop  24  is not reset. 
     Although the ON-state of the main switching transistor Tr 1  is detected from the gate-source voltage of the main switching transistor Tr 1  in each of the power supply control ICs  110 ,  130 ,  140 , and  150 , the ON-state of the main switching transistor Tr 1  may be detected from the output voltage Vout. 
     FIG. 16 is a block diagram of a fifth embodiment of the power supply control IC of the present invention. In this figure, the same components as in FIG. 12 are denoted by the same reference numerals, and explanations for those components are omitted in the following description. 
     A power supply control IC  170  differs from the power supply control IC  140  of FIG. 12 in that the flip-flop  24  is not reset. 
     As described so far, in accordance with the present invention, the output current of a DC-DC converter can be measured using the ON-resistance of the main switching transistor Tr 1 , instead of the current sense resistor. In the present invention, the gate-source voltage Vgs of the main switching transistor Tr 1  is monitored. When the gate-source voltage is higher than a predetermined value, the drain-source voltage of the main switching transistor Tr 1  is measured so as to measure the current flowing through the main switching transistor Tr 1 . Accordingly, the drain-source voltage of the main switching transistor Tr 1  is measured while the main switching transistor Tr 1  is certainly in the ON state. Thus, a wrong detection can be avoided. 
     The present invention is not limited to the specifically disclosed embodiments, but variations and modifications may be made without departing from the scope of the present invention. 
     The present application is based on Japanese priority application No. 2000-076994, filed on Mar. 17, 2000, the entire contents of which are hereby incorporated by reference.