Abstract:
A DC power source apparatus has a DC power source for supplying a DC voltage, a transformer, and a switching element connected to a primary winding of the transformer. The switching element carries out ON/OFF operations to convert the DC voltage from the DC power source into high-frequency power, which is transferred to a secondary winding of the transformer and is converted into a DC output voltage. The primary winding of the transformer consists of a first primary winding made of a plurality of winding layers and a second primary winding made of a plurality of winding layers. The first and second primary windings are connected in parallel. The first primary winding is arranged on an inner side of the secondary winding, and the second primary winding is arranged on an outer side of the secondary winding. A terminal of a winding layer farthest from the secondary winding among the winding layers of each of the first and second primary windings is connected to the switching element.

Description:
BACKGROUND OF THE INVENTION  
       [0001]     1. Field of the Invention  
         [0002]     The present invention relates to a DC power source apparatus, and particularly, to a power converting transformer for a DC power source apparatus.  
         [0003]     2. Description of the Related Art  
         [0004]      FIG. 1  is a circuit diagram showing a DC power source apparatus according to a related art. In the apparatus of  FIG. 1 , a DC power source E supplies a DC voltage. A switching element Q 1  is, for example, a MOSFET and carries out ON/OFF operations to convert the DC voltage from the DC power source E into high-frequency power. A transformer  1   a  is connected between the switching element Q 1  and a positive electrode of the DC power source E, so that the high-frequency power is transmitted from the primary side of the transformer  1   a  to the secondary side thereof. The high-frequency power on the secondary side of the transformer  1   a  is converted into a DC output voltage through a rectifying/smoothing circuit consisting of a diode D 1  and a smoothing capacitor C 1 . The DC output voltage is supplied to a load. An output voltage detector  3  detects the DC output voltage, compares the detected voltage with a reference voltage, and provides an error signal representative of the result of the comparison. According to the error signal, a control circuit (controller)  5  controls ON/OFF intervals of the switching element Q 1  so that a predetermined output voltage is supplied to the load.  
         [0005]     A tertiary winding D of the transformer  1   a  induces a voltage, which is rectified and smoothed through a diode D 2  and a capacitor C 2 . The rectified and smoothed voltage is supplied as a source voltage to the controller  5 .  
         [0006]      FIG. 2  is a sectional view showing the transformer  1   a  in the DC power source apparatus of  FIG. 1 ,  FIG. 3  is a view showing windings of the transformer  1   a,  and  FIG. 4  is a sectional view showing the transformer  1   a  and parasitic capacitance formed among the windings of the transformer  1   a.    
         [0007]     In  FIG. 2 , the transformer  1   a  has a core  11  made of magnetic material inserted into a bobbin  13 . Inside the bobbin  13 , first primary winding P 1 , a secondary winding S, a second primary winding P 2 , and the tertiary winding D are sequentially are sequentially arranged. The first primary winding P 1  consists of two winding layers P 1 - 1  and P 1 - 2 . The second primary winding P 2  consists of two winding layers P 2 - 1  and P 2 - 2 .  
         [0008]     Forming of the windings in the bobbin  13  will be explained. A wire is wound from a right end of the bobbin  13  in a vertical downward direction to form the winding layer P 1 - 1 . The wire is turned at a left end of the bobbin  13  and is wound to form the winding layer P 1 - 2  on the winding layer P 1 - 1 , thereby completing the first primary winding P 1 . On the winding layer P 1 - 2 , the secondary winding S is wound. Thereafter, the winding layers P 2 - 1  and P 2 - 2  are wound in the same direction as the winding layers P 1 - 1  and P 1 - 2 .  
         [0009]     To improve manufacturability, the windings of the transformer la are usually wound in the same direction. In  FIG. 1 , the first and second primary windings P 1  and P 2  are connected in parallel. In  FIGS. 2 and 4 , the secondary winding S is arranged between the first and second primary windings P 1  and P 2 , to increase the degree of coupling of these windings P 1 , P 2 , and S. In this case, there is parasitic capacitance C 112  between the winding layers P 1 - 1  and P 1 - 2 , parasitic capacitance C 12 S between the winding layer P 1 - 2  and the secondary winding S, parasitic capacitance C 21 S between the secondary winding S and the winding layer P 2 - 1 , and parasitic capacitance C 212  between the winding layers P 2 - 1  and P 2 - 2  produced.  
         [0010]     In  FIGS. 1 and 2 , the winding layer P 1 - 1  of the first primary winding P 1  and the winding layer P 2 - 1  of the second primary winding P 2  adjacent to the secondary winding S are on the switching element Q 1  side.  
         [0011]     The switching element Q 1  is continuously turned on and off therefore, the potential thereof greatly varies for the ON/OFF operations. The potential variations of the switching element Q 1  are applied to the first and second primary windings P 1  and P 2  of the transformer  1   a.  As a result, high-frequency currents pass through the parasitic capacitance C 12 S between the winding layer P 1 - 2  of the first primary winding P 1  and the secondary winding S and the parasitic capacitance C 21 S between the winding layer P 2 - 1  of the second primary winding P 2  and the secondary winding S to the secondary side of the transformer  1   a.    
         [0012]     Such high-frequency currents pass through a loop consisting of the first and second primary windings P 1  and P 2 , the secondary winding S, a circuitry on the secondary side, the ground, the parasitic capacitance between the ground and a circuitry on the primary side, the circuitry on the primary side, and the first and second primary windings P 1  and P 2 . Passing to the ground, the high-frequency currents cause common-mode noise. The common-mode noise leaks to the DC power source side and is radiated into space to badly affect other devices.  
         [0013]     When the switching element Q 1  is turned on, the DC voltage from the DC power source E is applied to a negative side of the first and second primary windings P 1  and P 2  of the transformer  1   a.  When the switching element Q 1  is turned off, a flyback voltage occurs on a positive side of the first and second primary windings P 1  and P 2 . Namely, first terminals of the first and second primary windings P 1  and P 2  connected to the switching element Q 1  are subjected to large potential variations, and second terminals thereof connected to the DC input voltage that is stable are subjected to no potential variation.  
         [0014]     The parasitic capacitance between the first and second primary windings P 1  and P 2  and the secondary winding S increases as the distance between them shortens. Accordingly, the high-frequency currents passing through the parasitic capacitance between the first and second primary windings P 1  and P 2  and the secondary winding Swill be large if the first terminals of the first and second primary windings P 1  and P 2  connected to the switching element Q 1  are close to the secondary winding S.  
         [0015]     In  FIG. 4 , the start of the first and second primary windings P 1  and P 2  are connected to the switching element Q 1 . The winding layer P 1 - 1  that is at the start of the first primary winding P 1  is located away from the secondary winding S, and the winding layer P 2 - 1  that is at the start of the second primary winding P 2  is located adjacent to the secondary winding S. Accordingly, a large high-frequency current passes the second primary winding P 2  through the parasitic capacitance C 21 S to the secondary winding S. In  FIGS. 1 and 4 , an arrow represents a high-frequency current with the width of the arrow indicating the magnitude of the current.  
         [0016]     To reduce the common-mode noise caused by high-frequency currents, FIGS.  5  to  7  show a transformer  1   b  according to another related art.  FIG. 5  is a sectional view showing the structure of the transformer  1   b,    FIG. 6  is a sectional view showing parasitic capacitance among windings of the transformer  1   b,  and  FIG. 7  is a circuit diagram showing a DC power source apparatus employing the transformer  1   b.    
         [0017]     The transformer  1   b  shown in FIGS.  5  to  7  has a shield plate  17  between a winding layer P 2 - 1  of a second primary winding P 2  and a secondary winding S, to reduce parasitic capacitance C 21 S between the winding layer P 2 - 1  and the secondary winding S. Reducing the parasitic capacitance C 21 S results in reducing a high-frequency current passing from the primary side of the transformer to the secondary side thereof, thereby decreasing the common-mode noise.  
       SUMMARY OF THE INVENTION  
       [0018]     The shield plate  17  of  FIG. 5  used to reduce the parasitic capacitance of the first and second primary windings complicates the structure of the transformer  1   b  and increases the distance between the primary and secondary windings by the thickness of the shield plate  17 . This results in increasing leakage inductance between the primary and secondary windings. Accordingly, the related art of  FIG. 5  is inappropriate to form a partial voltage resonance converter that uses a flyback voltage produced on a secondary winding to realize a zero-cross switch (zero-voltage switch ZVS).  
         [0019]     According to the present invention, a DC power source apparatus having a transformer that is structurally simple and is capable of reducing common-mode noise without increasing leakage inductance can be provided.  
         [0020]     In order to accomplish the objective, a first aspect of the present invention provides a DC power source apparatus having a transformer, a switching element connected to a primary winding of the transformer and configured to carry out ON/OFF operations to convert a DC voltage provided by a DC power source into high-frequency power, a rectifying/smoothing circuit configured to convert the high-frequency power transmitted from the primary winding to a secondary winding of the transformer into a DC output voltage, and a controller configured to control ON/OFF intervals of the switching element according to the DC output voltage so that a predetermined output voltage is supplied to a load in which the primary winding of the transformer includes a first primary winding having a plurality of winding layers and a second primary winding having a plurality of winding layers, the first primary winding is arranged on an inner side of the secondary winding, the second primary winding is arranged on an outer side of the secondary winding, and a terminal of a winding layer farthest from the secondary winding among the winding layers of each of the first and second primary windings is connected to the switching element.  
         [0021]     According to a second aspect of the present invention, the winding direction of the second primary winding is opposite to the winding direction of the first primary winding.  
         [0022]     According to a third aspect of the present invention, a terminal of a winding layer nearest to the secondary winding among the winding layers of each of the first and second primary windings is connected to the DC power source. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0023]      FIG. 1  is a circuit diagram showing a DC power source apparatus having a transformer according to a related art;  
         [0024]      FIG. 2  is a sectional view showing the structure of the transformer of  FIG. 1 ;  
         [0025]      FIG. 3  is a view showing windings of the transformer of  FIG. 1 ;  
         [0026]      FIG. 4  is a sectional view showing parasitic capacitance among the windings of the transformer of  FIG. 1 ;  
         [0027]      FIG. 5  is a sectional view showing the structure of a transformer according to another related art;  
         [0028]      FIG. 6  is a sectional view showing parasitic capacitance among the windings of the transformer of  FIG. 5 ;  
         [0029]      FIG. 7  is a circuit diagram showing a DC power source apparatus employing the transformer of  FIG. 5 ;  
         [0030]      FIG. 8  is a sectional view showing a transformer provided for a DC power source apparatus according to an embodiment of the present invention;  
         [0031]      FIG. 9  is a view showing windings of the transformer of  FIG. 8 ;  
         [0032]      FIG. 10  is a sectional view showing parasitic capacitance existing among the windings of the transformer of  FIG. 8 ;  
         [0033]      FIG. 11  is a circuit diagram showing a DC power source apparatus employing the transformer of  FIG. 8 ;  
         [0034]      FIG. 12  is a view showing voltages generated on the windings of the transformer when a switching element shown in  FIG. 11  is turned off;  
         [0035]      FIG. 13A  is a view showing voltages on the windings of the transformer according to the related art; and  
         [0036]      FIG. 13B  is a view showing voltages on the windings of the transformer according to the embodiment of the present invention. 
     
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT  
       [0037]     A DC power source apparatus according to an embodiment of the present invention will be explained in detail with reference to the accompanying drawings.  
         [0038]      FIG. 8  is a sectional view showing a transformer arranged in the DC power source apparatus according to the embodiment.  FIG. 9  is a view showing windings of the transformer of  FIG. 8 .  FIG. 10  is a sectional view showing parasitic capacitance among the windings of the transformer of  FIG. 8 .  FIG. 11  is a circuit diagram showing the DC power source apparatus employing the transformer of  FIG. 8 .  
         [0039]     In  FIG. 8 , the transformer  1  has a core  11  made of magnetic material. The core  11  is inserted into a bobbin  13 . Inside the bobbin  13 , a first primary winding P 1 , a secondary winding S, a second primary winding P 2   a,  and a tertiary winding D are sequentially arranged. The first primary winding P 1  has winding layers P 1 - 1  and P 1 - 2 . The second primary winding P 2   a  has winding layers P 2 - 2  and P 2 - 1 .  
         [0040]     Forming of the windings in the bobbin  13  will be explained. A wire is wound from a right end of the bobbin  13  in a vertical downward direction to form the winding layer P 1 - 1 . The wire is turned at a left end of the bobbin  13  and is wound to form the winding layer P 1 - 2  on the winding layer P 1 - 1 , thereby forming the first primary winding P 1 . On the winding layer P 1 - 2 , the secondary winding S is wound. These processes are the same as those of the related art.  
         [0041]     On the secondary winding S, a wire is wound in a vertical upward direction of the bobbin  13  to form the winding layer P 2 - 2 . The wire is turned at a left end of the bobbin  13  and is wound to form the winding layer P 2 - 1  on the winding layer P 2 - 2 , thereby forming the second primary winding P 2   a.    
         [0042]     In  FIG. 11 , the first primary winding P 1  and second primary winding P 2   a  are connected in parallel. In  FIG. 10 , there is parasitic capacitance C 112  existing between the winding layer P 1 - 1  and the winding layer P 1 - 2 , parasitic capacitance C 12 S existing between the winding layer P 1 - 2  and the secondary winding S, parasitic capacitance C 22 S existing between the secondary winding S and the winding layer P 2 - 2 , and parasitic capacitance C 212  existing between the winding layer P 2 - 1  and the winding layer P 2 - 2 .  
         [0043]     In  FIGS. 10 and 11 , the start (depicted with a large black dot) of the winding layer P 1 - 1  and the end of the winding layer P 2 - 1  are connected to a switching element Q 1 . The end of the winding layer P 1 - 2  and the start (depicted with a large black dot) of the winding layer P 2 - 2  are connected to a stable potential, i.e., a DC input voltage (on the positive electrode side of a DC power source E).  
         [0044]     The first primary winding P 1  and second primary winding P 2   a  are wound in the counter direction, and therefore, generate magnetic flux of the same phase when a current passes through the windings. Namely, the first and second primary windings P 1  and P 2   a  are connected in parallel. According to the embodiment, the start position and winding direction of the second primary winding P 2   a  are reversed in such a way that the stable potential side thereof is adjacent to the secondary winding S.  
         [0045]     According to the transformer  1  shown in  FIG. 8 , the winding layers P 1 - 1  and P 2 - 1  are arranged farthest from the secondary winding S, to reduce parasitic capacitance with respect to the secondary winding S because these winding layers P 1 - 1  and P 2 - 1  are connected to the switching element Q 1 , and therefore, are subjected to large voltage variations. The winding layers P 2 - 2  and P 1 - 2  are arranged nearest to the secondary winding S so that parasitic capacitance is large because the winding layers P 2 - 2  and P 1 - 2  are connected to the stable potential, and therefore, are subjected to small voltage variations. These configurations reduce high-frequency currents passing through the secondary winding S induced by current passing through the first and second primary windings P 1  and P 2   a.    
         [0046]     As is apparent in  FIGS. 10 and 11 , a high-frequency current passing from the winding layer P 2 - 1  to the secondary winding S is remarkably reduced. Accordingly, the transformer  1  of the embodiment is appropriate to form a partial voltage resonance converter that is structurally simple, is capable of reducing common-mode noise, and realizes a zero-cross switch with the use of a flyback voltage induced on the secondary winding S.  
         [0047]     Terminals of the winding layers P 1 - 2  and P 2 - 2  are connected to the positive electrode side of the DC power source E, and therefore, are subjected to no voltage variation. The winding layers P 1 - 2  and P 2 - 2  also serve as electromagnetic shields for the winding layers P 1 - 1  and P 2 - 1 , so that substantially no high-frequency current passes from the winding layers P 1 - 1  and P 2 - 1  to the secondary winding S. This further reduces common-mode noise.  
         [0048]     A main cause of the common-mode noise is a current passing from each of the first and second primary windings P 1  and P 2   a  to the secondary winding S. This current will be explained in detail.  
         [0049]     A capacitor is an element to accumulate charge. The quantity of charge in a capacitor is expressed with the capacitance C (corresponding to the parasitic capacitance of the embodiment) of the capacitor and a voltage V applied to the capacitor: 
 
Q=CV   (1) 
 
         [0050]     A current I corresponds to the quantity of charge that moves per unit time: 
 
 I=dQ/dt    (2) 
 
         [0051]     By differentiating the expression (1), the expressions (1) and (2) provide the following: 
 
 dQ/dt=I=CdV/dt    (3) 
 
         [0052]     According to the expression (3), it is understood that the current I is proportional to the capacitance and a voltage change per unit of time. Namely, a current passing from any one of the first and second primary windings P 1  and P 2   a  to the secondary winding S is proportional to the corresponding parasitic capacitance and a voltage change.  
         [0053]     The transformers of  FIGS. 4 and 10  substantially provide the same parasitic capacitance between the adjacent winding layers if the winding width of each winding layer is the same. Accordingly, in the transformer  1   a  of the related art, the parasitic capacitance between the winding layer P 1 - 2  and the secondary winding S that are close to each other and the parasitic capacitance between the winding layer P 2 - 1  and the secondary winding S that are close to each other is large. Also in the transformer  1   a,  the parasitic capacitance between the winding layer P 1 - 1  and the secondary winding S that are distant from each other and the parasitic capacitance between the winding layer P 2 - 2  and the secondary winding S that are distant from each other is small.  
         [0054]     According to the embodiment, a voltage variation on the winding layers P 1 - 2  and P 2 - 2  connected to the stable potential side is half a voltage variation on the winding layers P 1 - 1  and P 2 - 1  being connected to the switching element Q 1 , when measured from the stable potential side.  
         [0055]      FIG. 12  shows voltages generated on the windings of the transformer  1  when the switching element Q 1  is turned off. When the first primary winding P 1  induces a voltage Vp, each of the winding layers P 1 -l and P 1 - 2  induces a voltage of Vp/2 that is half the voltage Vp.  
         [0056]      FIG. 13A  shows voltages on the windings of the transformer la according to the related art of  FIGS. 1 and 2 , and  FIG. 13B  shows voltages on the windings of the transformer  1  according to the embodiment of the present invention. In  FIGS. 13A and 13B , a value between parentheses is a voltage induced on a winding with the stable potential side being a reference (zero).  
         [0057]     In  FIG. 13A , the transformer  1   a  of the related art shown in  FIGS. 1 and 2  induces a voltage of 0 to Vp/2 on the winding layer P 1 - 2  adjacent to the secondary winding S and a voltage of Vp/2 to Vp on the winding layer P 2 - 1  adjacent to the secondary winding S.  
         [0058]     On the other hand, the second primary winding P 2   a  is reversely arranged with respect to the first primary winding P 1  of the embodiment as shown in  FIGS. 8 and 11 . As a result, as shown in  FIG. 13B , each of the winding layers P 1 - 2  and P 2 - 2  that are adjacent to the secondary winding-S induces a voltage of 0 to Vp/2.  
         [0059]     Comparing the transformer  1  of the embodiment with the transformer  1   a  of the related art, it should be noted that, although the winding layer P 1 - 2  of each example induces the same voltage of 0 to Vp/2, the winding layer P 2 - 2  of the embodiment induces the voltage of 0 to Vp/2 that is half the voltage of Vp/2 to Vp induced by the winding layer P 2 - 1  of the related art that is arranged at the-same location as the winding layer P 2 - 2  of the embodiment. As is apparent in  FIGS. 10 and 11 , a high-frequency current passing from the winding layer P 2 - 2  to the secondary side through the parasitic capacitance C 22 S according to the embodiment is half a high-frequency current of the transformer  1   a  of the related art. Consequently, the transformer  1  of the embodiment can reduce common-mode noise.  
         [0060]     The second primary winding P 2   a  of the embodiment is reversed relative to the first primary winding P 1  without increasing the length of leads. Therefore, the winding layers P 1 - 2  and P 2 - 2  being adjacent to the secondary winding S can be connected to the stable potential side without increasing leakage inductance. As a result, the embodiment can reduce common-mode noise.  
         [0061]     According to the embodiment, each of the first and second primary windings P 1  and P 2   a  has two winding layers. Instead, each of the first and second primary windings P 1  and P 2   a  may have three or more winding layers.  
         [0062]     For example, in a case where each of the first and second primary windings P 1  and P 2   a  may be made of three winding layers, a terminal of a winding layer being farthest from the secondary winding S among the three layers in each of the first and second primary windings P 1  and P 2   a  is connected to the switching element Q 1 . And a terminal of a winding layer being nearest to the secondary winding S among the three layers in each of the first and second primary windings P 1  and P 2   a  is connected to the positive electrode side of the DC power source E.  
         [0063]     The winding direction of the second primary winding P 2   a  is made opposite to the winding direction of the first primary winding P 1 . In this case, a terminal voltage of the three winding layers in each of the first and second primary windings P 1  and P 2   a  is Vp. Then, a voltage variation on the winding layer nearest to the secondary winding S is reduced to 0 to Vp/3. This results in further reducing high-frequency currents passing from the first and second primary windings P 1  and P 2   a  to the secondary winding S. In this way, the three-layer winding can reduce high-frequency currents, i.e., common-mode noise more than the two-layer winding.  
       EFFECT OF THE INVENTION  
       [0064]     According to the first aspect of the present invention, the first primary winding is arranged on an inner side of the secondary winding and the second primary winding on an outer side of the secondary winding and a terminal of the winding layer farthest from the secondary winding among the winding layers of each of the first and second primary windings is connected to the switching element. Namely, each winding that is subjected to a large voltage variation is arranged at a location where parasitic capacitance is small, to reduce high-frequency currents passing from the first and second primary windings to the secondary winding. This configuration is appropriate to form a partial voltage resonance converter having a simple structure and capable of realizing a zero-cross switch with the use of a flyback voltage generated on the secondary winding.  
         [0065]     According to the second aspect of the present invention, the winding direction of the second primary winding is opposite to the winding direction of the first primary winding. If a terminal voltage of the winding layers of each of the first and second primary windings is Vp, a voltage variation on a winding layer nearest to the secondary winding is 0 to Vp/{the number of winding layers}. Due to such a small voltage variation, high-frequency currents passing from the first and second primary windings to the secondary winding are small. This results in further reducing common-mode noise. The second aspect of the present invention involves no increase in the number of leads, and therefore, never increases leakage inductance.  
         [0066]     According to the third aspect of the present invention, a terminal of a winding layer nearest to the secondary winding among the winding layers of each of the first and second primary windings is connected to the DC power source. As a result, the winding layer nearest to the secondary winding involves no voltage variation, and therefore, serves as a shield for a winding layer farthest from the secondary winding, the farthest winding layer being subjected to a large voltage variation. Consequently, there will be no high-frequency currents passing from the farthest winding layers to the secondary winding.  
       INDUSTRIAL APPLICABILITY  
       [0067]     The present invention is applicable to power source apparatuses such as DC-DC converters and AC-DC converters.  
         [0068]     This application claims benefit of priority under 35 USC §119 to Japanese Patent Applications No. 2004-249949, filed on Aug. 30, 2004, the entire contents of which are incorporated by reference herein. Although the invention has been described above by reference to certain embodiments of the invention, the invention is not limited to the embodiments described above. Modifications and variations of the embodiments described above will occur to those skilled in the art, in light of the teachings. The scope of the invention is defined with reference to the following claims.