Abstract:
An integrated circuit includes a filter circuit that has at least one active device, wherein the active device has adjustable transconductance.

Description:
BACKGROUND 
   In electronic devices such as semiconductor devices, in particular in corresponding analog and/or digital integrated circuits, memory devices, signal-processing systems etc., filter circuit arrangements, such as high and/or low pass filter circuit arrangements are frequently used. 
   A high pass is a filter circuit arrangement that transmits signals with relatively high frequencies substantially without modification. Further, with signals with relatively low frequencies in general attenuation and, as a rule, a phase lead is effected. 
   As compared to this, a low pass is a filter circuit arrangement in which signals with relatively low frequencies are transmitted substantially without modification; with signals with relatively high frequencies, attenuation and, as a rule, a phase lag is effected. 
   Passive high and/or low pass filter circuit arrangements may include one or several resistors and, as energy storage device, one or more capacitive devices and/or one or more inductive devices. 
   As compared to this, active filter circuit arrangements generally include, in addition to passive devices such as resistors and capacitors, one or more active devices, such as operational amplifiers. 
   For configuring the properties of a filter circuit arrangement such as the respective cutoff frequency, etc. switchable capacitors may be used. Alternatively or additionally, for configuring the properties of a filter circuit arrangement, the transconductance of active devices, e.g., transistors in respective operational amplifiers, etc. may be suitably adjusted, etc. However, changing a respective filter configuration due to relatively long transient times of control loops included in the respective filter circuit arrangement may last relatively long. For these or other reasons, there is a need for the present invention. 
   SUMMARY 
   According to an embodiment of the invention, a filter circuit arrangement is provided including at least one active device with adjustable transconductance. 
   Further features and advantages of the present invention will become apparent from the following detailed description of the invention made with reference to the accompanying drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The accompanying drawings are included to provide a further understanding of the present invention and are incorporated in and constitute a part of this specification. The drawings illustrate the embodiments of the present invention and together with the description serve to explain the principles of the invention. Other embodiments of the present invention and many of the intended advantages of the present invention will be readily appreciated as they become better understood by reference to the following detailed description. The elements of the drawings are not necessarily to scale relative to each other. Like reference numerals designate corresponding similar parts. 
       FIG. 1  illustrates a schematic, exemplary representation of a filter circuit arrangement in accordance with an embodiment of the invention. 
       FIG. 2  illustrates a schematic, exemplary representation of a filter circuit arrangement in accordance with an alternative embodiment of the invention. 
       FIG. 3  illustrates a schematic, exemplary representation of a portion of a filter circuit arrangement in accordance with an alternative embodiment of the arrangement illustrated in  FIG. 1 . 
   

   DETAILED DESCRIPTION 
   In the following Detailed Description, reference is made to the accompanying drawings, which form a part hereof, and in which is shown by way of illustration specific embodiments in which the invention may be practiced. In this regard, directional terminology, such as “top,” “bottom,” “front,” “back,” “leading,” “trailing,” etc., is used with reference to the orientation of the Figure(s) being described. Because components of embodiments of the present invention can be positioned in a number of different orientations, the directional terminology is used for purposes of illustration and is in no way limiting. It is to be understood that other embodiments may be utilized and structural or logical changes may be made without departing from the scope of the present invention. The following detailed description, therefore, is not to be taken in a limiting sense, and the scope of the present invention is defined by the appended claims. 
     FIG. 1  illustrates a schematic, exemplary representation of a filter circuit arrangement  100  in accordance with an embodiment of the invention. 
   As results from  FIG. 1 , the filter circuit arrangement  100  includes a plurality of n-channel field effect transistors  101 ,  102 ,  103 ,  104 ,  105 , and optionally,  201 ,  202 ,  203 ,  204 . 
   Further, the filter circuit arrangement  100  includes several p-channel field effect transistors  227 ,  228 , a plurality of capacitors  254 ,  206 ,  302 , and a linear amplifier device  301 . 
   In addition, and as will be described in further detail below, the filter circuit arrangement  100  includes a plurality of switches  401 ,  402 ,  403 ,  501 ,  502 ,  503 . 
   As is illustrated in  FIG. 1 , the drain of the n-channel field effect transistor  102  is connected to the drain of the n-channel field effect transistor  103 . Further, the drain of the n-channel field effect transistor  102  and the drain of the n-channel field effect transistor  103  are connected via a line  601  with the drain of the p-channel field effect transistor  227 , with the source of the p-channel field effect transistor  228 , and with a current input line  602 . Further, the source of the p-channel field effect transistor  227  is connected to a supply voltage. 
   The drain of the p-channel field effect transistor  228  is connected with the drain of the n-channel field effect transistor  105 , the source of which is connected to ground. 
   Further, the drain of the n-channel field effect transistor  105  is connected with the gate of the n-channel field effect transistor  201 , the gate of the n-channel field effect transistor  101 , and the capacitor  254 , which is also connected to ground. 
   As is also illustrated in  FIG. 1 , the source of the n-channel field effect transistor  102  is connected with the drain of the n-channel field effect transistor  101 , the capacitor  302 , the first connection of the switch  403 , and the capacitor  206 , which also is connected to ground. 
   Further, the source of the n-channel field effect transistor  103  is connected with the drain of the n-channel field effect transistor  104 , and to the second connection of the switch  403 . 
   The gate of the n-channel field effect transistor  104  is connected to the first connection of the switch  401 , and to the first connection of the switch  402 . Further, the source of the n-channel field effect transistor  104  is connected to ground. Correspondingly similar, the source of the n-channel field effect transistor  101  is connected to ground. 
   The second connection of the switch  402  is connected to the gate of the n-channel field effect transistor  101 , and hence, also to the capacitor  254 , the drains of the field effect transistors  105 ,  228 , and the gate of the field effect transistor  201 . 
   In addition, the second connection of the switch  401  is biased to a (constant or substantially constant) voltage U_ref_gain by a voltage source  252  (see remarks below). 
   Further, the capacitor  302  is connected to an output of the linear amplifier device  301 , the input of which is connected to a line  603 . 
   As is further illustrated in  FIG. 1 , the gate of the n-channel field effect transistor  103  is connected to the gate of the n-channel field effect transistor  102 , which via a line  604  is connected to the gate of the n-channel field effect transistor  202 , and to the gate of the n-channel field effect transistor  203 . 
   In addition, the line  604  is biased to a (constant) voltage U_refc 1  by using a voltage source  253 . Furthermore, the gate of the p-channel field effect transistor  228  is biased to a (constant) voltage U_refc 2  by using a voltage source  251 . 
   As is also illustrated in  FIG. 1 , the source of the n-channel field effect transistor  202  is connected with the drain of the n-channel field effect transistor  201 , and to the first connection of the switch  503 . 
   Further, the source of the n-channel field effect transistor  203  is connected with the drain of the n-channel field effect transistor  204 , and to the second connection of the switch  503 . 
   The gate of the n-channel field effect transistor  204  is connected to the first connection of the switch  501 , and to the first connection of the switch  502 . Further, the source of the n-channel field effect transistor  204  is connected to ground. Correspondingly similar, the source of the n-channel field effect transistor  201  is connected to ground. 
   The second connection of the switch  501  is connected to the gate of the n-channel field effect transistor  201 , and hence, also to the capacitor  254 , the drains of the field effect transistors  105 ,  228 , and the gate of the field effect transistor  101 . 
   In addition, the second connection of the switch  502  is biased to the above (constant or substantially constant) voltage U_ref_gain by a voltage source  255  (see remark below). 
   As is further illustrated in  FIG. 1 , the drain of the n-channel field effect transistor  202  is connected to the drain of the n-channel field effect transistor  203 . Further, the drain of the n-channel field effect transistor  202  and the drain of the n-channel field effect transistor  203  are connected with a voltage output line  605 , and via a load resistor  250  to the above supply voltage. 
   As indicated in  FIG. 1 , by use of the amplifier device  301 , a voltage U_in present at the line  603 , i.e., the voltage U_in present at the input of the amplifier device  301  is amplified by the amplifier device  301  by a respective amplifying factor k, such that a voltage of k×U_in is put out at its output, and hence, is fed to the capacitor  302 . 
   Further, in the present embodiment, the voltage U_in present at the line  603 , i.e., the voltage U_in representing the input of the amplifier device  301  is chosen to be R 1  times I_in, i.e., R 1  times an input signal current I_in present at the above current input line  602  (such that U_in=I_in×R 1 ). 
   As is illustrated in  FIG. 1 , by use of the filter circuit arrangement  100 , the input current I_in fed to the filter circuit arrangement  100  via the current input line  602 , and representing the input of the filter circuit arrangement  100  is converted into a respective output voltage U_out that can be tapped at the above voltage output line  605 , and representing the output of the filter circuit arrangement  100 . 
   In the filter circuit arrangement  100 , the p-channel field effect transistor  228  serves as a cascode to keep the node connected with the line  601 /the drains of the n-channel field effect transistors  102 ,  103  on a predefined potential. 
   Further, the p-channel field effect transistor  227  acts as a current source (current IQ 2 ). Such a current source may in alternative embodiments be designed differently than illustrated in  FIG. 1 . 
   As results further from  FIG. 1 , the n-channel field effect transistor  105  acts as a current sink (current IQ 1 ). Hence, the pairs of n-channel field effect transistors  102 ,  103  or  101 ,  104 , respectively are thus flown through by a current that results from the difference between the above-mentioned currents IQ 2  and IQ 1  (IQ 2 −IQ 1 ). A current sink as provided by the transistor  105  may in alternative embodiments be designed differently than illustrated in  FIG. 1 . 
   As becomes clear from what was said above, the above transistors  201 ,  202 ,  203 ,  204  e.g., serve to couple out respective signals from the circuit arrangement  100 , in particular, e.g., the above output voltage U_out. 
   The circuit arrangement  100  may be operated in several different frequency bands. 
   If the circuit arrangement  100  is to be operated in a first frequency band including relatively high frequencies (“high frequency band”), the (total/effective) transconductance of active devices, e.g., transistors in the circuit arrangement  100  may be adjusted, e.g., may be made higher. 
   For this purpose, e.g., the switches  402 ,  403 ,  501 ,  503  are closed, and the switches  401 ,  502  are opened. 
   In this case, the n-channel field effect transistors  101 ,  104 , as well as the n-channel field effect transistors  102 ,  103 , as well as the n-channel field effect transistors  201 ,  204 , as well as the n-channel field effect transistors  202 ,  203  are connected in parallel, respectively. 
   In this case, and as will be described in further detail below, the n-channel field effect transistor  101  together with the n-channel field effect transistor  104  has a (total/effective) transconductance of gm 1 . Correspondingly similar, also, the n-channel field effect transistor  201  together with the n-channel field effect transistor  204  has a (total/effective) transconductance of gm 1 . Further, the n-channel field effect transistor  102  together with the n-channel field effect transistor  103  has a (total/effective) transconductance of gm 2 . Correspondingly similar, also, the n-channel field effect transistor  202  together with the n-channel field effect transistor  203  has a (total/effective) transconductance of gm 2 . 
   The current control loop e.g., including the transistor  101  in parallel to the transistor  104 , the transistor  102  in parallel to the transistor  103 , the above cascode transistor  228 , the above current sink, and the above current source controls the current through the transistor  101  in parallel to the transistor  104 , and the transistor  102  in parallel to the transistor  103  so that the above current IQ 2 −IQ 1  flows through the pairs of transistors  102 ,  103  and  101 ,  104 , respectively. 
   Correspondingly similar, and by a corresponding current control loop, the current which flows through the above pairs of transistors  202 ,  203  and  201 ,  204 , respectively, which serve to couple out respective signals from the circuit arrangement  100 , also is controlled such that it is IQ 2 −IQ 1 . 
   In an exemplary embodiment, the above (constant or substantially constant) biasing voltage U_ref_gain (voltage sources  252  and  255 , respectively) may be provided by a (circuit) configuration working at the same or substantially the same current level as the above current control loops, but through which no signal current (such as the above input signal current I_in) flows. 
   Hence, it is ensured that the biasing voltage U_ref_gain is the same or substantially the same as a voltage at a node  703  (“gain” voltage) (i.e., the voltage at the drain of the transistor  105 , the drain of the transistor  228 , and the gates of the transistors  101 ,  201 ) if no input signal current I_in is fed to the circuit arrangement  100 . Further, it is ensured that the transistors  101  and  104 , the transistors  102  and  103 , the transistors  201  and  204 , and the transistors  202  and  203  have the same/a constant current density, even when switching between respective frequency bands (see below). 
   The transconductance of a field effect transistor, in particular, a MOS field effect transistor is as follows (equation 1): 
   
     
       
         
           
             g 
             m 
           
           = 
           
             
               2 
               ⁢ 
               
                   
               
               ⁢ 
               μ 
               ⁢ 
               
                   
               
               ⁢ 
               
                 C 
                 ox 
               
               ⁢ 
               
                 W 
                 L 
               
               ⁢ 
               
                 I 
                 D 
               
             
           
         
       
     
   
   Hence, as follows from this equation, due to the above same/constant current density through the transistors  101 ,  104 , and  102 ,  103 , and  201 ,  204 , and  202 ,  203 , respectively, by changing the transistor width W, the transconductance gm may be linearly controlled, as by changing the transistor width W, also, the drain current I D  is changed. A condition for the above-said is that the transistors all have the same or substantially the same length L, which generally is the case. 
   As an example, and as is illustrated in  FIG. 1 , the transistor  101  alone works at e.g., a transconductance of n×gm 1 . Together/in parallel with the transistor  104 , the transistors  101 ,  104  work at the above (total/effective) transconductance of gm 1 , as the transistor  104  alone as also illustrated in  FIG. 1  works at e.g., a transconductance of (1−n)×gm 1 , and the ratio between the width of the transistor  101  and the total width of the transistors  101 ,  104  is n (with n&lt;1). For instance, the width of the transistor  101  may be 3 μm, and the width of the transistor  104  may be 7 μm, and the total width of the transistors  101 ,  104  may hence be 10 μm. In this case, the transistor  101  alone works at e.g., a transconductance of 30% of the (total/effective) transconductance of the transistors  101 ,  104  working together/in parallel. 
   The same applies to the above transistors  201 ,  204 , whereby as also illustrated in  FIG. 1 , the transistor  201  alone also works at e.g., the above transconductance of n×gm 1 , and the transistor  204  alone also works at e.g., a transconductance of (1−n)×gm 1 , such that (also) the transistors  201  and  204  together/in parallel work at the above (total) transconductance of gm 1 . 
   Further, and as also illustrated in  FIG. 1 , the transistor  102  alone works at e.g., a transconductance of n×gm 2 . Hence, together/in parallel with the transistor  103 , the transistors  102 ,  103  work at the above (total) transconductance of gm 2 , as the transistor  103  alone as also illustrated in  FIG. 1  works at e.g., a transconductance of (1−n)×gm 2 . 
   The same e.g., applies to the above transistors  202 ,  203 , whereby as also illustrated in  FIG. 1 , the transistor  202  alone works at e.g., the above transconductance of n×gm 2 , and the transistor  203  alone works at e.g., a transconductance of (1−n)×gm 2 , such that (also) the transistors  202  and  203  together/in parallel work at the above (total) transconductance of gm 2 . 
   If the circuit arrangement  100  is to be operated in a second frequency band different from the above first frequency band, and including relatively low frequencies (“low frequency band”), the (total/effective) transconductance of active devices, e.g., transistors in the circuit arrangement  100  may be adjusted, e.g., may be made lower. 
   For this purpose, e.g., the switches  402 ,  403 ,  501 ,  503  are opened, and the switches  401 ,  502  are closed. 
   Hence, the gate of the n-channel field effect transistor  104  is connected to the above (constant or substantially constant) biasing voltage U_ref_gain (voltage source  252 ). Correspondingly similar, the gate of the n-channel field effect transistor  204  is connected to the above (constant or substantially constant) biasing voltage U_ref_gain (voltage source  255 ). 
   As said above, the biasing voltage U_ref_gain is the same or substantial the same as the dc voltage at the node  703  (“gain” voltage). 
   Hence, the transistors  104 ,  204  represent a current bank with constant current density above the transistors; the level of the dc voltage at the node  703  (“gain” voltage) is not influenced by the above switching (i.e., the switching from the above “high frequency band” to the above “low frequency band”, and vice versa, by the above opening/closing of the switches  402 ,  403 ,  501 ,  503  and  401 ,  502 , as explained above). 
   After switching to the above “low frequency band” operation, the same dc current flows through the transistors  104 ,  103 ,  204 ,  203  as before, i.e., during the operation of the circuit arrangement  100  in the above “high frequency band”, but without a signal current (i.e., a respective ac current, in particular, e.g., the above input signal current I_in). 
   Further, after switching to the above “low frequency band” operation, the same dc current flows through the transistors  101 ,  102 ,  201 ,  202  as before, i.e., during the operation of the circuit arrangement  100  in the above “high frequency band”, but in addition, again (as in the “high frequency band”), the signal current (i.e., a respective ac current, in particular, e.g., the above input signal current I_in). 
   However, the transistors  101 ,  102 ,  201 ,  202  as becomes clear from what was explained above now operate at a reduced (effective) transconductance, here: at a transconductance e.g., reduced by the above factor n. 
   For example, the transistors  101 ,  201  during the above “low frequency band” operation e.g., work at a transconductance of n×gm 1  (instead of at an effective transconductance of gm 1  as during the above “high frequency band” operation). 
   Further, the transistors  102 ,  202  during the above “low frequency band” operation e.g., work at a transconductance of n×gm 2  (instead of at an effective transconductance of gm 2  as during the above “high frequency band” operation). 
   By the circuit arrangement  100 , a low pass is formed having a complex-conjugated double pole, and an adjustable (positive or negative) zero point. 
   The resonance frequency of the double pole results as follows (for the “high frequency band” operation): 
             f   0     =       1     2   ⁢           ⁢   π       ⁢         gm   ⁢           ⁢   1   ⁢           ⁢   gm   ⁢           ⁢   2       C   ⁢           ⁢   1   ⁢     (       C   ⁢           ⁢   2     +     C   ⁢           ⁢   3       )                   
and the attenuation (also for the “high frequency band” operation) as follows:
 
           ζ   =       1   2     ⁢         gm   ⁢           ⁢   2   ⁢           ⁢   C   ⁢           ⁢   1       gm   ⁢           ⁢   1   ⁢     (       C   ⁢           ⁢   2     +     C   ⁢           ⁢   3       )                   
Further, the positive/negative zero point results as follows (for the “high frequency band” operation):
 
   
     
       
         
           
             f 
             0 
           
           = 
           
             
               gm 
               ⁢ 
               
                   
               
               ⁢ 
               2 
             
             
               2 
               ⁢ 
               
                   
               
               ⁢ 
               
                 π 
                 ⁡ 
                 
                   ( 
                   
                     
                       C 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       2 
                     
                     + 
                     
                       C 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       3 
                     
                     - 
                     
                       kR 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       1 
                       ⁢ 
                       gm 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       2 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       C 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       3 
                     
                   
                   ) 
                 
               
             
           
         
       
     
   
   Hence, by suitably choosing the above amplifying factor k of the amplifier device  301  to be positive or negative, the zero point frequency may be correspondingly chosen to be positive or negative, such that the total transfer function of the filter may be chosen such that the filter effects a respective phase lead (in the case of a positive zero point) or a respective a phase lag (in the case of a negative zero point). 
   In an alternative variant of the circuit arrangement  100  illustrated in  FIG. 1 , the transistors provided there may—instead as in the above-explained embodiment in NMOS or PMOS technology—e.g., also be designed in bipolar or BiCMOS technology. Alternatively, a complementary circuit implementation is, for instance, also conceivable, in which PMOS transistors are used instead of NMOS transistors, and vice versa NMOS transistors instead of PMOS transistors. 
     FIG. 2  illustrates a schematic, exemplary representation of a filter circuit arrangement  1000  in accordance with an alternative embodiment of the invention. 
   As results from  FIG. 2 , the filter circuit arrangement  1000  includes a plurality of n-channel field effect transistors  1101 ,  1102 ,  1103 ,  1104 ,  1105 ,  901 ,  903  and—optionally— 1201 ,  1202 ,  1203 ,  1204 . 
   Further, the filter circuit arrangement  1000  includes several p-channel field effect transistors  1227 ,  1228 , a plurality of capacitors  1254 ,  1206 ,  1302 , a plurality of resistors  904 ,  905 , and a linear amplifier device  1301 . 
   In addition, and as will be described in further detail below, the filter circuit arrangement  1000  includes a plurality of switches  1401 ,  1402 ,  1403 ,  1501 ,  1502 ,  1503 ,  902 ,  906 . 
   As is illustrated in  FIG. 2 , the drain of the n-channel field effect transistor  1102  is connected to the source of the n-channel field effect transistor  901 , and—via the resistor  904 —is connected to ground. 
   Further, the drain of the n-channel field effect transistor  1103  is connected to the source of the n-channel field effect transistor  903 , and—via the resistor  905 —is connected to ground. 
   Still further, the drain of the n-channel field effect transistor  901  and the drain of the n-channel field effect transistor  903  are connected with the drain of the p-channel field effect transistor  1227 , and with the source of the p-channel field effect transistor  1228 . 
   The gate of the n-channel field effect transistor  903  is connected to the first connection of the switch  902 , and to the first connection of the switch  906 , the second connection of which is biased to a voltage U_ref_cm by a voltage source  907 . 
   Further, the gate of the n-channel field effect transistor  901  is connected to the second connection of the switch  902 , and to a line  1603 , which is connected to an input of the amplifier device  1301 . 
   As is further illustrated in  FIG. 2 , the source of the p-channel field effect transistor  1227  is connected to a supply voltage. 
   The drain of the p-channel field effect transistor  1228  is connected with the drain of the n-channel field effect transistor  1105 , the source of which is connected to ground. 
   Further, the drain of the n-channel field effect transistor  1105  is connected with the gate of the n-channel field effect transistor  1201 , the gate of the n-channel field effect transistor  1101 , and the capacitor  1254 , which is also connected to ground. 
   As is also illustrated in  FIG. 2 , the source of the n-channel field effect transistor  1102  is connected with the drain of the n-channel field effect transistor  1101 , the capacitor  1302 , the first connection of the switch  1403 , and the capacitor  1206 , which also is connected to ground. 
   Further, the source of the n-channel field effect transistor  1103  is connected with the drain of the n-channel field effect transistor  1104 , and to the second connection of the switch  1403 . 
   The gate of the n-channel field effect transistor  1104  is connected to the first connection of the switch  1401 , and to the first connection of the switch  1402 . Further, the source of the n-channel field effect transistor  1104  is connected to ground. Correspondingly similar, the source of the n-channel field effect transistor  1101  is connected to ground. 
   The second connection of the switch  1402  is connected to the gate of the n-channel field effect transistor  1101 , and hence, also to the capacitor  1254 , the drains of the field effect transistors  1105 ,  1228 , and the gate of the field effect transistor  1201 . 
   In addition, the second connection of the switch  1401  is biased to a voltage U_ref_gain by a voltage source  1252 . 
   Further, the capacitor  1302  is connected to an output of the linear amplifier device  1301 , the input of which is connected to the line  1603  (here: a respective voltage input line  1603 ). 
   As is further illustrated in  FIG. 2 , the gate of the n-channel field effect transistor  1103  is connected to the gate of the n-channel field effect transistor  1102 , which via a line  1604  is connected to the gate of the n-channel field effect transistor  1202 , and to the gate of the n-channel field effect transistor  1203 . 
   In addition, the line  1604  is biased to a voltage U_refc 1  by a voltage source  1253 . Furthermore, the gate of the p-channel field effect transistor  1228  is biased to a voltage U_refc 2  by a voltage source  1251 . 
   As is also illustrated in  FIG. 2 , the source of the n-channel field effect transistor  1202  is connected with the drain of the n-channel field effect transistor  1201 , and to the first connection of the switch  1503 . 
   Further, the source of the n-channel field effect transistor  1203  is connected with the drain of the n-channel field effect transistor  1204 , and to the second connection of the switch  1503 . 
   The gate of the n-channel field effect transistor  1204  is connected to the first connection of the switch  1501 , and to the first connection of the switch  1502 . Further, the source of the n-channel field effect transistor  1204  is connected to ground. Correspondingly similar, the source of the n-channel field effect transistor  1201  is connected to ground. 
   The second connection of the switch  1501  is connected to the gate of the n-channel field effect transistor  1201 , and hence, also to the capacitor  1254 , the drains of the field effect transistors  1105 ,  1228 , and the gate of the field effect transistor  1101 . 
   In addition, the second connection of the switch  1502  is biased to the voltage U_ref_gain by a voltage source  1255 . 
   Further, the drain of the n-channel field effect transistor  1202  is connected to the drain of the n-channel field effect transistor  1203 . In addition, the drain of the n-channel field effect transistor  1202  and the drain of the n-channel field effect transistor  1203  are connected with a voltage output line  1605 , and via a load resistor  1250  to the above supply voltage. 
   As is illustrated in  FIG. 2 , by use of the filter circuit arrangement  1000 , an input voltage U_in fed to the filter circuit arrangement  1000  at the above voltage input line  1603 , and representing the input of the filter circuit arrangement  1000  is converted into a respective output voltage U_out that can be tapped at the above voltage output line  1605 , and representing the output of the filter circuit arrangement  1000 . 
   In the filter circuit arrangement  1000 , the p-channel field effect transistor  1228  serves as a cascode to keep the node connected with the drains of the n-channel field effect transistors  901 ,  903  on a predefined potential. 
   Further, the p-channel field effect transistor  1227  acts as a current source (current IQ 2 ). Such a current source may in alternative embodiments be designed differently than illustrated in  FIG. 2 . 
   As results further from  FIG. 2 , the n-channel field effect transistor  1105  acts as a current sink (current IQ 1 ). A current sink as provided by the transistor  1105  may in alternative embodiments be designed differently than illustrated in  FIG. 2 . 
   As becomes clear from what was said above, the above transistors  1201 ,  1202 ,  1203 ,  1204  e.g., serve to couple out respective signals from the circuit arrangement  1000 , in particular, e.g., the above output voltage U_out. 
   The circuit arrangement  1000  may be operated in several different frequency bands. 
   If the circuit arrangement  1000  is to be operated in a first frequency band including relatively high frequencies (“high frequency band”), the (total/effective) transconductance of active devices, e.g., transistors in the circuit arrangement  1000  may be adjusted, e.g., may be made higher. 
   For this purpose, e.g., the switches  1402 ,  1403 ,  902 ,  1501 ,  1503  are closed, and the switches  1401 ,  1502 ,  906  are opened. 
   In this case, the n-channel field effect transistors  1101 ,  1104 , as well as the n-channel field effect transistors  1102 ,  1103 , as well as the n-channel field effect transistors  1201 ,  1204 , the n-channel field effect transistors  1202 ,  1203 , as well as the n-channel field effect transistors  901 ,  903  are connected in parallel, respectively. 
   Hence, the n-channel field effect transistor  1101  together with the n-channel field effect transistor  1104  has a (total/effective) transconductance of gm 1 . Correspondingly similar, also, the n-channel field effect transistor  1201  together with the n-channel field effect transistor  1204  has a (total/effective) transconductance of gm 1 . Further, the n-channel field effect transistor  1102  together with the n-channel field effect transistor  1103  has a (total/effective) transconductance of gm 2 . Correspondingly similar, also, the n-channel field effect transistor  1202  together with the n-channel field effect transistor  1203  has a (total/effective) transconductance of gm 2 . Further, the n-channel field effect transistor  901  together with the n-channel field effect transistor  903  has a (total/effective) transconductance of gm 3 . 
   The current control loop e.g., including the transistor  1101  in parallel to the transistor  1104 , the transistor  1102  in parallel to the transistor  1103 , the transistor  901  in parallel to the transistor  903 , the resistor  905  (having a resistance of R 1 /(n−1)), the resistor  904  (having a resistance of R 1 /n), the above cascode transistor  1228 , the above current sink (current IQ 1 ), and the above current source (current IQ 2 ) controls the current through the transistor  1101  in parallel to the transistor  1104 , the transistor  1102  in parallel to the transistor  1103 , and the transistor  901  in parallel to the transistor  903  so that a current of IQ 2 −IQ 1  flows through the respective pairs of transistors  1102 ,  1103 ,  1101 ,  1104 ,  901 ,  903 . 
   Correspondingly similar, and by a corresponding current control loop, the current which flows through the above pairs of transistors  1202 ,  1203  and  1201 ,  1204 , respectively, which as said e.g., serve to couple out respective signals from the circuit arrangement  1000 , also is controlled such that it is IQ 2 −IQ 1 . 
   In an exemplary embodiment, the above (constant or substantially constant) biasing voltage U_ref_gain (voltage sources  1252  and  1255 , respectively) may be provided by a (circuit) configuration working at the same or substantially the same current level as the above current control loops, but through which no signal current flows. 
   Hence, it is ensured that the biasing voltage U_ref_gain is the same or substantial the same as a voltage at a node  1703  (“gain” voltage) (i.e., the voltage at the drain of the transistor  1105 , the drain of the transistor  1228 , and the gates of the transistors  1101 ,  1201 ) if no input signal (such as the above input voltage U_in) is fed to the circuit arrangement  1000 . Further, it is ensured that the transistors  1101  and  1104 , the transistors  1102  and  1103 , the transistors  1201  and  1204 , the transistors  1202  and  1203 , and the transistors  901 ,  903  have the same/a constant current density, even when switching between respective frequency bands (see below). 
   The transistor  1101  alone works at e.g., a transconductance of n×gm 1 . Together/in parallel with the transistor  1104 , the transistors  1101 ,  1104  work at a (total/effective) transconductance of gm 1 , as the transistor  1104  alone as illustrated in  FIG. 2  works at e.g., a transconductance of (1−n)×gm 1 , and the ratio between the width of the transistor  1101  and the total width of the transistors  1101 ,  1104  is n (with n&lt;1). For instance, the width of the transistor  1101  may be 3 μm, and the width of the transistor  1104  may be 7 μm, and the total width of the transistors  1101 ,  1104  may hence be 10 μm. In this case, the transistor  1101  alone works at e.g., a transconductance of 30% of the (total/effective) transconductance of the transistors  1101 ,  1104  working together/in parallel. 
   The same e.g., applies to the above transistors  1201 ,  1204 , as also illustrated in  FIG. 2 , the transistor  1201  alone also works at e.g., the above transconductance of n×gm 1 , and the transistor  1204  alone also works at e.g., a transconductance of (1−n)×gm 1 , such that (also) the transistors  1201  and  1204  together/in parallel work at the above (total) transconductance of gm 1 . 
   Further, and as also illustrated in  FIG. 2 , the transistor  1102  alone works at e.g., a transconductance of n×gm 2 . Hence, together/in parallel with the transistor  1103 , the transistors  1102 ,  1103  work at a (total) transconductance of gm 2 , as the transistor  1103  alone as also illustrated in  FIG. 2  works at e.g., a transconductance of (1−n)×gm 2 . 
   The same e.g., applies to the above transistors  1202 ,  1203 , whereby as also illustrated in  FIG. 2 , the transistor  1202  alone works at e.g., the above transconductance of n×gm 2 , and the transistor  1203  alone works at e.g., a transconductance of (1−n)×gm 2 , such that (also) the transistors  1202  and  1203  together/in parallel work at the above (total) transconductance of gm 2 . 
   Still further, and as also illustrated in  FIG. 2 , the transistor  901  alone works at e.g., a transconductance of n×gm 3 . Hence, together/in parallel with the transistor  903 , the transistors  901 ,  903  work at a (total) transconductance of gm 3 , as the transistor  903  alone as also illustrated in  FIG. 3  works at e.g., a transconductance of (1−n)×gm 3 . 
   If the circuit arrangement  1000  is to be operated in a second frequency band different from the above first frequency band, and including relatively low frequencies (“low frequency band”), the (total/effective) transconductance of active devices, e.g., transistors in the circuit arrangement  1000  may be adjusted, e.g., may be made lower. 
   For this purpose, e.g., the switches  1402 ,  1403 ,  902 ,  1501 ,  1503  are opened, and the switches  1401 ,  1502 ,  906  are closed. 
   Hence, the gate of the n-channel field effect transistor  1104  is connected to the above (constant or substantially constant) biasing voltage U_ref_gain (voltage source  1252 ). Correspondingly similar, the gate of the n-channel field effect transistor  1204  is connected to the above (constant or substantially constant) biasing voltage U_ref_gain (voltage source  1255 ). 
   As said above, the biasing voltage U_ref_gain is the same or substantial the same as the dc voltage at the node  1703  (“gain” voltage). 
   Hence, the transistors  1104 ,  1204  represent a current bank with constant current density above the transistors; the level of the dc voltage at the node  1703  (“gain” voltage) is not influenced by the above switching (i.e., the switching from the above “high frequency band” to the above “low frequency band”, and vice versa, by the above opening/closing of the switches  1402 ,  1403 ,  902 ,  1501 ,  1503  and  1401 ,  1502 ,  906  as explained above). 
   After switching to the above “low frequency band” operation, the same dc current flows through the transistors  1104 ,  1103 ,  903 ,  1204 ,  1203  as before, i.e., during the operation of the circuit arrangement  1000  in the above “high frequency band”, but without a signal current. 
   Further, after switching to the above “low frequency band” operation, the same dc current flows through the transistors  1101 ,  1102 ,  901 ,  1201 ,  1202  as before, i.e., during the operation of the circuit arrangement  1000  in the above “high frequency band”, but in addition, again (as in the “high frequency band”), the signal current. 
   However, the transistors  1101 ,  1102 ,  901 ,  1201 ,  1202  as becomes clear from what was explained above now operate at a reduced (effective) transconductance, here: at a transconductance e.g., reduced by the above factor n. 
   For example, the transistors  1101 ,  1201  during the above “low frequency band” operation e.g., work at a transconductance of n×gm 1  (instead of at an effective transconductance of gm 1  as during the above “high frequency band” operation). 
   Further, the transistors  1102 ,  1202  during the above “low frequency band” operation e.g., work at a transconductance of n×gm 2  (instead of at an effective transconductance of gm 2  as during the above “high frequency band” operation). 
   Still further, the transistor  901  during the above “low frequency band” operation e.g., works at a transconductance of n×gm 3  (instead of at an effective transconductance of gm 3  as during the above “high frequency band” operation). 
   The voltage U_ref_cm provided by the voltage source  907  corresponds to the (static) dc voltage at the gate of the transistor  901  when no input signal is present. Hence, by the resistors  904 ,  905  the respective dc current is correctly adjusted, such that regardless of the state of the circuit arrangement  1000  (“high frequency band” operation, or “low frequency band” operation) the same or substantially the same voltage potential is present at the sources of the transistors  901 ,  903 . 
   In order not to change the dynamic properties of the circuit arrangement  1000  whilst switching between the above frequency bands, also the resistors  904 ,  905  at the sources of the transistors  901 ,  903  are dimensioned according to the respective transistor transconductances. Hence, the resistance of the resistor  905  is chosen to be R 1 /(1−n), and the resistance of the resistor  904  is chosen to be R 1 /n. During the above “high frequency band” operation, the resistors  904 ,  905  are connected in parallel, such that the (total/effective) resistance of the resistors  904 ,  905  then is R 1 . 
   The circuit arrangement  1000  illustrated in  FIG. 2  (and also the circuit arrangement  100  illustrated in  FIG. 1 ) may according to further alternative variants be constructed “differentially”. In this case, e.g., the resistors  904 ,  905  may not be connected to ground. Instead, the resistors  904 ,  905  may then e.g., be connected with each other. 
   By the circuit arrangement  1000 , a low pass is formed having a complex-conjugated double pole, and an adjustable (positive or negative) zero point. 
   The resonance frequency of the double pole results as follows (for the “high frequency band” operation): 
             f   0     =       1     2   ⁢           ⁢   π       ⁢         gm   ⁢           ⁢   1   ⁢           ⁢   gm   ⁢           ⁢   2   ⁢           ⁢   gm   ⁢           ⁢   3   ⁢           ⁢   R   ⁢           ⁢   1       C   ⁢           ⁢   1   ⁢     (       C   ⁢           ⁢   2     +     C   ⁢           ⁢   3       )     ⁢     (     1   +     gm   ⁢           ⁢   3   ⁢           ⁢   R   ⁢           ⁢   1       )                   
and the attenuation (also for the “high frequency band” operation) as follows:
 
           ζ   =       1   2     ⁢         gm   ⁢           ⁢   2   ⁢   C   ⁢           ⁢   1   ⁢     (     1   +     gm   ⁢           ⁢   3   ⁢           ⁢   R   ⁢           ⁢   1       )         gm   ⁢           ⁢   1   ⁢     (       C   ⁢           ⁢   2     +     C   ⁢           ⁢   3       )     ⁢   gm   ⁢           ⁢   3   ⁢           ⁢   R   ⁢           ⁢   1                 
Further, the positive/negative zero point results as follows (for the “high frequency band” operation):
 
   
     
       
         
           
             f 
             0 
           
           = 
           
             
               gm 
               ⁢ 
               
                   
               
               ⁢ 
               2 
             
             
               2 
               ⁢ 
               
                   
               
               ⁢ 
               
                 π 
                 ⁡ 
                 
                   ( 
                   
                     
                       C 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       2 
                     
                     + 
                     
                       C 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       3 
                     
                     - 
                     
                       kR 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       1 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       gm 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       2 
                       ⁢ 
                       C 
                       ⁢ 
                       
                           
                       
                       ⁢ 
                       3 
                     
                   
                   ) 
                 
               
             
           
         
       
     
   
   Hence, by suitably choosing the above amplifying factor k of the amplifier device  1301  to be positive or negative, the zero point frequency may be correspondingly chosen to be positive or negative, such that the total transfer function of the filter may be chosen such that the filter effects a respective phase lead (in the case of a positive zero point) or a respective a phase lag (in the case of a negative zero point). 
   In an alternative variant of the circuit arrangement  1000  illustrated in  FIG. 2 , the transistors provided there may—instead as in the above-explained embodiment in NMOS or PMOS technology—e.g., also be designed in bipolar or BiCMOS technology. Alternatively, a complementary circuit implementation is, for instance, also conceivable, in which PMOS transistors are used instead of NMOS transistors, and vice versa NMOS transistors instead of PMOS transistors. 
   In further, alternative variants, a plurality (e.g., two, three, or more) of the filter circuit arrangements  100 ,  1000  illustrated in  FIGS. 1 and 2  may be connected in series (cascading). 
     FIG. 3  illustrates a schematic, exemplary representation of a portion of a filter circuit arrangement in accordance with an alternative embodiment of the arrangement  100  illustrated in  FIG. 1 . 
   As results from  FIG. 3 , instead of the pair of n-channel field effect transistors  101 ,  104  as illustrated in  FIG. 1 , a first set of three transistors (three n-channel field effect transistors  2101 ,  2104 ,  2104   a ) is used. In addition, instead of the pair of n-channel field effect transistors  102 ,  103 , a second set of three transistors (n-channel field effect transistors  2102 ,  2103 ,  2103   a ) are used. 
   In addition, and as will be described in further detail below, instead of the switches  401 ,  402 ,  403 , the switches  2401 ,  2402 ,  2403 ,  2401   a ,  2402   a ,  2403   a  are used. 
   As is illustrated in  FIG. 3 , a first connection of the first switch  2402  is connected to a control gate of a first transistor  2104  of the first set of transistors, a second connection of the first switch  2402  is connected to a control gate of a second transistor  2101  of the first set of transistors. Further, a first connection of the second switch  2401  is connected to the control gate of the first transistor  2104  of the first set of transistors, and a second connection of the second switch  2401  is connected to a reference voltage U_ref_gain. In addition, a first connection of the third switch  2402   a  is connected to a control gate of a third transistor  2104   a  of the first set of transistors, and a second connection of the third switch  2402   a  is connected to the control gate of the second transistor  2101  of the first set of transistors. A first connection of the fourth switch  2401   a  is connected to the control gate of the third transistor  2104   a  of the first set of transistors, and a second connection of the fourth switch  2401   a  is connected to a reference voltage U_ref_gain. 
   In addition, and as is further illustrated in  FIG. 3 , a first connection of the fifth switch  2403  is connected to the first transistor  2104  of the first set of transistors and to a first transistor  2103  of the second set of transistors, and a second connection of the fifth switch  2403  is connected to the second transistor  2101  of the first set of transistors and to a second transistor  2102  of the second set of transistors. Further, a first connection of the sixth switch  2403   a  is connected to the third transistor  2104   a  of the first set of transistors and to a third transistor  2103   a  of the second set of transistors, and a second connection of the sixth switch  2403   a  is connected to the second transistor  2101  of the first set of transistors and to the second transistor  2102  of the second set of transistors. 
   The circuit arrangement may be operated in three different frequency bands. If the circuit arrangement is to be operated in a first frequency band including very high frequencies (“very high frequency band”), the (total/effective) transconductance of active devices, e.g., transistors in the circuit arrangement may be adjusted, e.g., may be high. For this purpose, e.g., the switches  2402 ,  2402   a ,  2403 ,  2403   a  are closed, and the switches  2401 ,  2401   a  are opened. If the circuit arrangement is to be operated in a second frequency band including high frequencies (“high frequency band”), but frequencies lower than in the “very high frequency band”, the (total/effective) transconductance of active devices, e.g., transistors in the circuit arrangement may be adjusted, e.g., may be made lower. For this purpose, e.g., the switches  2402 ,  2401   a ,  2403  are closed, and the switches  2401 ,  2402   a ,  2403   a  are opened. If the circuit arrangement is to be operated in a third frequency band including relatively low frequencies (“low frequency band”), i.e., frequencies lower than in the “high frequency band”, the (total/effective) transconductance of active devices, e.g., transistors in the circuit arrangement may again be adjusted, e.g., may be made even lower. For this purpose, e.g., the switches  2401 ,  2401   a  are closed, and the switches  2402 ,  2402   a ,  2403 ,  2403   a  are opened. 
   Although specific embodiments have been illustrated and described herein, it will be appreciated by those of ordinary skill in the art that a variety of alternate and/or equivalent implementations may be substituted for the specific embodiments illustrated and described without departing from the scope of the present invention. This application is intended to cover any adaptations or variations of the specific embodiments discussed herein. Therefore, it is intended that this invention be limited only by the claims and the equivalents thereof.