Abstract:
The invention is a digital sounder module and its method for detection. The digital sounder module includes a sonar carrier wave producing means, a band-pass filter, a pre-amplifier for providing a high sensitivity and extending a wide dynamic range, and an analog-to-digital converter for providing a digital implementation of a superheterodyne detector and producing an intermediate frequency. The digital sounder module also includes a programmable logic device for controlling a gain of the pre-amplifier and for digitally filtering the intermediate frequency and a microprocessor. The method implemented by the controlled per-amplifier gain processes the return echo signal by controlled ramp up of the gain over time to compensate for the change in signal strength in proportion to the length of the return path.

Description:
I claim the benefit under Title 35, United States Code, § 120 to U.S. Provisional Application No. 60/430,939, filed Dec. 5, 2002, entitled DIGITAL SOUNDER MODULE AND METHOD FOR DETECTING. 

   BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   This invention relates to a sonar system and a method for detecting with the sonar system. Specifically, the invention relates to a sonar system and a method for detecting for use as a recreational fish finder. 
   2. Description of the Related Art 
   A sonar system typically consists of a transmitter, a transducer, a receiver, and a display. The transmitter of the sonar system transmits an electrical impulse. The transducer converts the electrical impulse into a sound wave and sends the sound wave through the water. When the sound wave strikes an object, the sound wave rebounds and returns to the sonar system. The sonar return wave strikes the transducer of the sonar system and the transducer converts the sonar return wave into an electrical signal. The receiver filters, amplifies, and detects the sonar return signal. The receiver converts the sonar return signal into a digital signal after its detection and sends the amplified signal to the display. 
   The receiver section of a conventional sonar system comprises an analog filter and an analog detector. Such conventional sonar receiver sections can be built using a Philips SA604 “High Performance low power FM IF system.” The Philips SA604 is a monolithic low-power FM intermediate frequency (IF) system that incorporates two limiting IF amplifiers, a quadrature detector, muting, a logarithmic received signal strength indicator, and a voltage regulator. The Philips SA604 is typically used in radios for use as an IF amplifier. The Philips SA604 includes a Received Signal Strength Indicator (RSSI) for outputting a logarithmic representation of the signal strength on the IF input. 
   A disadvantage of conventional sonar systems is that they are manually tuned during the production and do allow for a later adjustment of the center frequency. As a result, tuning cannot be optimized for use with a specific transducer. 
   An additional disadvantage of conventional sonar systems is that the bandwidth is fixed. A fixed bandwidth makes it impossible for the sonar system to retrieve an optimal time/frequency response in both shallow and deep water, thus affecting the resolution. 
   Conventional sonar systems have a dynamic range of approximately 120 dB. An analog-to-digital (A/D) converter with a minimum resolution of 20 bits is required to sample sonar return waves digitally within this dynamic range. The expense of a commercially available A/D converter with 20 bits of resolution is too high for practical use in a recreational fishfinder. 
   It is an object of the invention to provide a sonar system that performs a digital filtration and a digital detection of a sonar carrier wave. 
   An additional object of the invention is to provide a sonar system with a high sensitivity and a wide dynamic range. 
   An additional object of the invention is to provide a sonar system with a programmable frequency. 
   Another object of the invention is to provide a sonar system with an automatic tuning that can be optimized for use with a specific transducer. 
   Another object of the invention is to provide a sonar system with a programmable bandwidth. 
   Another object of the invention is to provide a sonar system with an optimal time/frequency response in both shallow and deep water. 
   Another object of the invention is to provide a sonar receiver section with commercially available components that are of low cost and are suitable for use in a recreational fishfinder. 
   SUMMARY OF THE INVENTION 
   The problems and disadvantages of the conventional sonar systems discussed above are overcome, and other advantages are provided, by the following invention. The invention includes a sonar system with a high sensitivity, a wide dynamic range, a programmable frequency, and a programmable bandwidth. The invention also includes a method for performing a digital filtration and a digital detection of a sonar return wave. 
   The invention is a digital sounder module. The digital sounder module includes a sonar carrier wave producing means, a band-pass filter, a pre-amplifier for providing a high sensitivity and extending a wide dynamic range, and an analog-to-digital converter for providing a digital implementation of a superheterodyne detector and producing an intermediate frequency. The digital sounder module also includes a programmable logic device for controlling a gain of the pre-amplifier and for digitally filtering the intermediate frequency and a microprocessor. 
   The invention also relates to a method for performing a digital filtration and a digital detection of a sonar return wave. This method for detecting a sonar carrier wave includes receiving a sonar carrier wave, limiting the sonar carrier wave with a band-pass filter, and amplifying the filtered sonar carrier wave with a pre-amplifier. Then the method includes feeding the amplified sonar carrier wave to an analog-to-digital converter to produce an intermediate frequency of the sonar carrier wave. Next, controlling the pre-amplifier and the analog-to-digital converter with a programmable logic device is performed. This step is followed by filtering digitally the sonar carrier wave and detecting digitally the sonar carrier wave. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  illustrates a sonar system in accordance with the invention. 
       FIG. 2  illustrates a functional diagram of a digital sounder module in accordance with the invention. 
       FIG. 3  illustrates a harmonic sampling of the sonar return signal. 
       FIG. 4  illustrates an algorithm for approximating a phase shift. 
       FIG. 5  illustrates an algorithm for approximating the Euclidean distance between the 90 degree out-of-phase vectors. 
       FIG. 6  illustrates the logarithmic function implemented as a piecewise linear log 2 ( ) function. 
       FIG. 7  illustrates a transducer mounted on the hull of a vessel with echo reflections from three targets at varied depth. 
       FIGS. 8A through 8D  illustrates the modification of the echo return signal by the signal processing of the present invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   The invention includes a sonar system with a high sensitivity, a wide dynamic range, a programmable frequency, and a programmable bandwidth. The invention also includes a method for performing a digital filtration and a digital detection of a sonar return wave. 
     FIG. 1  illustrates a sonar system  100  in accordance with the invention. The sonar system  100  includes a sonar sound wave  101 , a sonar return wave  102 , a transducer  103 , a sonar return signal  104 , a digital sounder module (DSM)  150 , and a display  124 . The DSM  150  includes a band-pass filter  106 , a filtered return signal  108 , a pre-amplifier  110 , an amplified return signal  112 , an analog-to-digital (A/D) converter  114 , an intermediate frequency (IF) return signal  116 , a programmable logic device (PLD)  118 , a digitized return signal  120 , a microprocessor  122 , a compressed signal  123 , and a pre-amplifier gain curve  126 , illustrated in FIG.  8 C. 
   As illustrated in  FIG. 7 , the transducer  103  of the sonar system  100  mountedto the vessel  99 , transmits a sonar sound wave  101  out into the water. When the sonar sound wave  101  strikes an object, such as exemplary objects A, B and C, the sonar sound wave  101  rebounds and returns to the transducer  103  of the sonar system  100 . The sonar return waves  102  strike the transducer  103  of the sonar system  100 . 
   The transducer  103  can be either a thru-hull, an in-hull, or a transom mount transducer. The transducer  103  emits a sonar acoustic wave  101  at either 200 kHz or 50 kHz in the exemplary embodiments, however any sonar wave can perform in acordance with the teachings of the present invention. The transducer  103  receives the sonar return wave  102 , as a series of reflections of sonar wave  101  from objects in the water and from the bottom of the body of water if within range of the sonar detection capabilities. By measuring the return time of the echo, the distance traveled can be determined. The received echo is converted into an electrical sonar return signal  104 . The transducer  103  then sends the sonar return signal  104  to the DSM  150 . The general characteristics of a typical sonar echo return signal  104  with three echos are illustrated in extremely simplified form in FIG.  8 A. 
   The PLD controls the systems and adjusts such parameters as anticipated ping depth dependent upon return echo characteristics or user selected sonar parameters. In order to pass the initial unmodified signal with least clipping, first band pass filter  106  is set to the widest bandpass. The programable band pass filter which adjusts according to depth of application, is in the PLD. A narrow band pass filter is selected by the PLD to give good performance in deep water a wider bandpass is selected for shallow water. The transducer  103  pings are sent once every two seconds to twenty six times per second, less often in deeper water (once every two seconds) to allow for echo return time and more often (up to twenty six times per second) in shallow water. The PLD matches a narrow bandpass with less frequent pulses and a wider bandpass with a more frequent pulse rate. 
   The signal  104 ,  FIG. 8A , has a carrier frequency of 50 or 200 kHz depending upon the frequency of the transducer  103 . Regions of increased magnitude in the wave form (A, B and C) indicate echos of the signal reflected by objects in the water. Three echos are indicated for simplified illustrative purposes. A return sonar may have more or fewer echos. The position along the wave is indicative of the depth of the object. Echo A is a reflection from an object closer to the surface than object B, which in turn is closer than object C. The magnitude m and duration l of the region of increased magnitude is indicative of the size of the object causing the reflected echo. However, because of the characteristics of acoustic sonar echo reflection, the echo reflections of similarly sizes objects can vary significantly in magnitude depending upon the distance to the object. 
   The DSM  150  is the receiver of the sonar system  100 . A preferred embodiment of the DSM  150  is a digital implementation of a superheterodyne detector with a programmable frequency and a programmable bandwidth. The DSM  150  provides a programmable frequency range of 10 kHz to 200 KHz. The DSM  150  receives the sonar return signal  104  from the transducer  103  and the sonar return signal  104  is input to the band-pass filter  106  where it is band-pass limited. 
   The band-pass filter  106  is a typical wide-banded band-pass filter. The band-pass filter  106  is centered around 50 kHz or 200 kHz (depending upon the frequency of the signal from the transducer  1  as determined by the PLD  118 , based upon water conditions such as depth) in an preferred embodiment. The selection of frequency is controlled by the PLD  118 , based upon various factors, as described below. This can easily be modified to filter any frequency between 10 kHz and 200 kHz. The band-pass filter  106  supplies a filtered signal,  8 B to pre-amplifier  110 . The band-pass filter  106  sends the filtered return signal  108  to the pre-amplifier  110  for modification by the pre-amp according to a variable gain. 
   The pre-amplifier  110  is a two-stage differential variable gain amplifier using general purpose operational amplifiers. The gain of the amplifier is set by the gain curve  126 . The pre-amplifier  110  provides the DSM  150  with a high sensitivity and a wide dynamic range. The high sensitivity and the wide dynamic range of the pre-amplifier  110  allow for the use of a low cost A/D converter with a lower resolution while still maintaining the required dynamic range of the DSM  150 . The pre-amplifier  110  is digitally controlled by the PLD  118 . The pre-amplifier  110  amplifies the filtered return signal  108  based upon the pre-amplifier gain curve  126  that is sent by the PLD  118 . 
   The gain curve  126 ,  FIG. 8C , begins at a minimum gain level and increases the gain provided to amplifier  110  over time. In this manner, echos from greater depths receive significantly greater gain than echos from shallow depths. The shape of the gain curve, an important feature of the present invention, is adopted to reduce the significant difference in magnitude between signals from shallow and deep echos. Because sonar is significantly attenuated in water, the deeper the echo reflection source the greater the attenuation. By providing a gain which increases with time, the difference in echo magnitude can be partially compensated. The magnitude difference illustrated in  FIG. 8A  has been partially compensated as illustrated in FIG.  8 D. The pre-amplifier  110  outputs the compensated return signal  112 ,  FIG. 8D , to the A/D converter  114 . 
   The pre-amplifier gain curve  126  increases with depth and allows for a continuous reprogramming of a programmable filter, which provides an optimal performance at any depth/pulse width. The pre-amplifier gain curve  126  provides a sensitivity time control function that reduces surface clutter and background noise by decreasing the gain in shallow water. 
   The A/D converter  114  is an analog-to-digital converter. A preferred embodiment of the A/D converter  114  is an LTC® 1861 by Linear Technology Corporation. The LTC® 1861 is a 12 bit A/D converter that offers a software-selectable two-channel multiplexer. The amplified carrier wave is converted to digital by an A/D converter. The rate at which the carrier wave is converted to digital acts as a harmonic sub-sampling and the carrier wave is transformed into an IF frequency when digitized. The A/D converter  114  converts the amplified return signal  112  into an IF sonar return signal  116 . The A/D converter  114  outputs and sends the IF sonar return signal  116  to the PLD  118 . 
   The PLD  118  is a programmable logic device and operates as a co-processor for the main processor  122 . A preferred embodiment of the PLD  118  is an XILINX® Spartan IIe XC2S300E. The PLD  118  uses a simplified implementation of a noncoherent windowed in-phase/quadrature (IQ) detector to filter the IF sonar return signal  116  digitally. The method for digitally filtering the IF sonar return signal  116  is illustrated in  FIGS. 2 through 5  and discussed in further detail below. The PLD  118  outputs the pre-amplifier gain curve  126  to control the pre-amplifier  110  and outputs a digitized signal  120  to the microprocessor  122 . 
   The microprocessor  122  is illustrated in FIG.  6  and discussed in further detail below. 
   The display  124  receives the compressed signal  123  from the microprocessor  122  and outputs a display that may contain the water depth, temperature, and/or speed, as measured by the transducer  103 , and a dual or a split frequency fishfinder display, as detected by the DSM  150 . 
     FIG. 2  illustrates a functional diagram of a DSM  150  in accordance with the invention. The functional diagram of the DSM  150  includes the sonar return signal  104 , the band-pass filter  106 , the filtered return signal  108 , the pre-amplifier  110 , the amplified signal  112 , an A/D converter  114 , the IF sonar return signal  116 , the PLD  118 , and the digitized return signal  120 . The PLD  118  includes a window filter  206 , a windowed return signal  208 , a power measurement function  210 , a power signal  212 , a signal compression function  214 , the pre-amplifier gain curve  126 . 
   The digital signal in an exemplary embodiment, is filtered using a single point Fourier transformation and a programmable Kaiser window. The magnitude signal is converted to a logarithmic representation. 
   The sonar return signal  104  is band-pass limited by the band-pass filter  106 . The filtered return signal  108  is amplified by the pre-amplifier  110  based upon the pre-amplifier gain curve  126  as sent from the PLD  118 . The amplified return signal  112  is input into the A/D converter  114 , where it is mixed with a cosφ signal sent from the PLD  118  and converted into the IF sonar return signal  116 . The PLD  118  sets the sampling rate to insure that the sonar return signal is above the Nyquist frequency. The effect of this sub-sampling is a frequency shift which is mathematically described as “mixing with a cosφ signal.” 
   The PLD  118  detects the IF sonar return signal  116  by use of a simplified implementation of a noncoherent windowed in-phase/quadrature (I/Q) detector. The PLD  118  performs a Fourier transform of the IF sonar return signal  116 . The Fourier transform converts the IF sonar return signal  116  into a complex return signal which includes a real value that is an in-phase detected signal and an imaginary value that is a quadrature detected signal. The PLD  118  then multiplies the complex return signal by the window filter  206 . 
   The window filter  206  sets the bandwidth and the frequency roll-off. A preferred embodiment of the window filter  206  is a Kaiser window. 
   The power measurement function  210  measures the sonar return signal  104  by taking the absolute value of the windowed return signal  208  at a set frequency point. 
   The signal compression function  214  compresses the dynamic range of the DSM  150  by taking the log 2  of the power signal  212  and outputs the digitized return signal  120 . 
   The center frequency of the DSM  150  can be altered by adjusting the k value of the window filter  206 . 
   The time/frequency response of the DSM  150  can be altered by adjusting the window function w[n] of the window filter  206 . 
     FIGS. 3A and 3B  illustrates a harmonic sampling of the sonar return signal. The DSM  150  samples the sonar carrier wave through the use of harmonic sampling. The DSM  150  “undersamples” sonar carrier waves containing frequencies above the Nyquist sampling rate (one-half the sampling frequency, f s ) as shown in FIG.  3 A. Such undersampling of the sonar return signal is possible because the sonar carrier wave is a baseband signal. Sonar carrier waves containing frequencies below the Nyquist rate are directly sampled, as shown in FIG.  3 B. The A/D converter  114  samples the amplified return signal  112  and converts the amplified return signal  112  to an IF sonar return signal  116 . 
     FIG. 4  illustrates an algorithm for approximating a phase shift. It is not necessary to calculate both the in-phase and the quadrature component of the IF sonar return signal  116 . An approximation can instead be performed by approximating either component as a 90 degree phase shift of the other. Such an approximation reduces the complexity of the calculations by half with only a few resulting minor errors. The PLD  118  approximates this phase shift by taking a Fourier transform  400  of the Σw[n]e (jnθ)/k  signal from the window filter  206  (shown in FIG.  2 ), where sign(x) equals 1 if x&gt;0, 0 if x=0, and −1 if x&lt;0. The square of an in-phase detected signal  402  is taken by a square function  404 . A delay element  415  produces a pseudo Quadrature signal  416  of the in-phase detected signal  402 . The square of the pseudo Quadrature signal  416  is then taken by a square function  417 . The Euclidian distance  412  is calculated by taking the square root  410  of the sum  406  of the squared in-phase signal  405  and the squared Quadrature signal  418 . The digitized sonar return signal  120  is then compressed and output by the log 2  function  414 . 
     FIG. 5  illustrates an algorithm for approximating the Euclidian distance between 90 degree out-of-phase vectors. This approximation also provides a significant decrease in complexity without any significant sampling loss of the sonar return signal. 
   The filtered digital signal is compared with the previous result, and an approximated magnitude is calculated where the magnitude=max(y[n],y[n−1])+0.5*min(y[n],y[n−1]). This magnitude signal is converted to logarithmic representation through an approximated log2(x) calculation. The approximated log 2  is a piecewise linear approximation of a log 2  function where the log value is linear approximated between all integer log 2  values. 
   The PLD  118  approximates the Euclidian distance  506  by first taking a Fourier transform  400  of the Σw[n]e (jnθ)/k  signal from the window filter  206  (shown in FIG.  2 ), where sign(x) equals 1 if x&gt;0, 0 if x=0, and −1 if x&lt;0. The length of the longest in-phase detected signal  502  is then added to half of the length of the shortest in-phase detected signal  503  to produce the approximation of the Euclidian distance  506 . The digitized sonar return signal  120  is then compressed and output by the log 2  function  414 . 
     FIG. 6  illustrates the logarithmic function implemented as a piecewise linear log 2 ( ) function. A left-shift of the magnitude signal from the digital sonar module is performed until a “1” appears in the most significant bit. The signal is then sent through a multiplexer and a register.