Abstract:
A switching power source device for supplying power to a load includes a series resonant circuit, a plurality of main switch elements or main switch element groups for switching a current path of the series resonant circuit, a transformer for inducing a secondary current from the series resonant circuit, a plurality of synchronous rectification switch elements for rectifying the secondary current, a maximum on width control circuit for ordering a start and a completion of a maximum on width to the synchronous rectification switch element in synchronization with a timing of turning on the main switch elements or the main switch element groups, and a synchronous control circuit. The circuit controls an on period of the synchronous rectification switch element so as to turn on the synchronous rectification switch element in synchronization with a particular timing, and turn off in synchronization with another timing.

Description:
BACKGROUND OF THE INVENTION AND RELATED ART STATEMENT 
     The present invention relates to a switching power source device, a switching power source control circuit, and a switching power source device control method, including a series resonant circuit which has a current resonant inductor and a current resonant capacitor, and in particular relates to a switching power source device, a switching power source control circuit, and a switching power source device control method which eliminate a reverse flow of a current at a time of a light load. 
     As an existing switching power source device, one which includes the kind of current resonant type converter is known ( FIG. 5 ). In the current resonant type converter, an input direct voltage Vi is applied to a series resonant circuit which includes a resonant inductor Lr and a resonant capacitor Cr. By turning on and off two main switch elements Qa, Qb configured of metal-oxide-semiconductor field-effect transistors (MOSFET), or the like, and controlling a path of a primary side current flowing in a first coil L 1  of a power converting transformer T, a current with a sinusoidal waveform flows in the first coil L 1  of the transformer T. Also, rectification diodes D 1 , D 2 , which rectify induced secondary currents I 1 , I 2  respectively, and an output capacitor C 0 , which smoothes an output voltage V 0  to a load LD, are connected to a second coil L 2  and a third coil L 3  of the transformer T (a coil ratio of L 1 :L 2 :L 3  is taken to be n:1:1). Furthermore, the output voltage V 0  to the load LD is returned to a drive circuit  3  of the main switch elements Qa, Qb via an error amplifier  1  and a voltage control oscillation circuit (VCO)  2 , controlling a current and voltage flowing in the first coil L 1  of the transformer T, and controlling the output voltage V 0  at a constant voltage. The VCO  2  functions so that, when determining that due to an output of the error amplifier  1 , the output voltage V 0  is higher than a set voltage, or is a light load, it increases an output frequency thereof, while when determining that the output voltage V 0  is lower than the set voltage, or is a heavy load, it reduces the output frequency thereof. 
     However, when using this kind of switching power source device as a low voltage with high current power source, when the secondary currents I 1 , I 2  flow into the rectification diodes D 1 , D 2  provided on the secondary side of the transformer T, due to a forward drop voltage V F  of the rectification diodes D 1 , D 2 , a large power loss V F ×I 0  occurs. I 0  is either one of the secondary currents I 1 , I 2 . Therefore, as shown in  FIG. 6 , a separately excited drive type of current resonant circuit is used in which MOSFETs Qs 1 , Qs 2 , each of which has low on resistance, are connected as synchronous rectification switch elements in place of the rectification diodes D 1 , D 2 , a synchronous rectification is carried out, and the power loss is reduced. The MOSFETs Qs 1 , Qs 2  shown in  FIG. 6  is individually on-off controlled by the drive circuit  3  in synchronization with an operation frequency fop for turning on and off the primary side main switch elements Qa, Qb, so that the secondary currents I 1 , I 2  are alternately accumulated in the capacitor C 0 . 
     Since this kind of switching power source device is configured in such a way as to cause a switching operation of the main switch elements Qa, Qb, and obtain an optional direct current output via the voltage converting transformer T. Therefore, a charge accumulated in the capacitor C 0  is discharged, a current flowing back to the transformer T side (a reverse current) occurs, and a power loss in a reverse flow area becomes a problem depending on a size of the load LD connected to the secondary side, and the like. 
       FIG. 7  is a circuit diagram illustrating one example of a secondary side diode rectification current resonant type converter. In  FIG. 7 , the transformer T of the current resonant type converter of  FIG. 5  is divided into an exciting inductance element Lm and an ideal transformer Ti, and an operating principle thereof is clearly illustrated. Herein, prior to a description of the power loss in the reverse flow area, a description will be given of the operating principle of the current resonant type converter. 
     With the kind of current resonant type converter shown in  FIG. 7 , it is possible to define the following two kinds of basic current resonant frequency. Herein, Lr, Lm, and Cr indicate an inductance of the resonant inductor Lr, the exciting inductance element of the transformer T, and a capacitance of the resonant capacitor Cr, respectively. 
     
       
         
           
             
               
                 
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                           Lr 
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                           Cr 
                         
                       
                     
                   
                 
               
               
                 
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                               Lr 
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                           Cr 
                         
                       
                     
                   
                 
               
               
                 
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     With the switching power source device of  FIGS. 5 and 7 , when there is a supply of power to the load LD, the voltage of the exciting inductance element Lm of the transformer T is clamped at n×(V 0 +V F ) in accordance with the output voltage V 0 , the exciting inductance element Lm is not involved in the current resonance, and power is supplied to the secondary side circuit by operating at a first resonant frequency fr 1  (refer to Expression 1 above) decided by the resonant capacitor Cr and resonant inductor Lr. In this case, a sum of a current Im and a resonant current Ir flowing in the exciting inductance element Lm flows as a charge-discharge current to the resonant capacitor Cr. At this time, the operation frequency fop of the main switch elements Qa, Qb is controlled by the VCO  2  so as to stabilize the output voltage V 0 . 
     A second resonant frequency fr 2  is a resonant frequency when no supply of power is carried out to the load LD connected to the secondary side of the transformer T. Since the ideal transformer Ti does not function as a transformer, and the voltage of the exciting inductance element Lm of the transformer T is not clamped, a resonance operation is carried out mainly by the capacitance Cr of the resonant capacitor Cr, the resonant inductance Lr of the resonant inductor Lr, and the exciting inductance element Lm. 
     Regarding a specific resonance operation of the current resonant type converter of  FIG. 7 , it is possible to divide into six operational modes (Modes  1  to  6 ), as shown in  FIG. 8 , depending on a relationship between its operation frequency fop and the first resonance frequency fr 1  (hereafter called simply the resonance frequency), and on the size of the load LD connected to the secondary side of the transformer T. 
     That is, in  FIG. 8 , Modes  1  to  3  are cases in which the operation frequency fop is lower than the resonance frequency fr 1 , while Modes  4  to  6  are cases in which the operation frequency fop is equal to or higher than the resonance frequency fr 1 . Also, a heavy load (HL) condition is such that the size of the connected load LD is 50% or more of a rated load (maximum load) of the switching power source device, a light load (LL) condition is such that the size is 20% to 50%, and a very light load (VLL) condition is such that the size is 20% or less. 
     Firstly, a description will be given for the operational modes of the current resonant type converter of  FIG. 5  by using the operation waveforms shown in  FIG. 9 . 
       FIG. 9  shows an operation waveform of the first operational mode (Mode  1 ), in which the operation frequency fop (one cycle of the switching operation is taken to be “Top”) is lower than the resonance frequently fr 1  (the resonance frequency at that time is taken to be “Tr”), and the load LD is in the heavy load condition. Herein, a description will be given for one operation frequency Top which is divided into ten operational conditions (timings s 1  to s 10 ), in order from the timing s 1  (hereafter, the term “timing sj” (j=1 to 16) indicates a time period, rather than an instant) immediately after one main switch element Qa is turned on. 
     In  FIG. 9 , (A) and (B) are gate voltages Vga, and Vgb of the main switch elements Qa, and Qb, which are outputs of the drive circuit  3 . C and D are currents Ia and Ib which flow in the main switch elements Qa, Qb. E is an inter-terminal voltage Vc of the resonant capacitor Cr, and F and G are currents I 1  and I 2  which flow in the rectification diodes D 1  and D 2 . 
       FIG. 10  is a circuit diagram illustrating the operational condition in the timing s 1  of the first operational mode. Herein, a state immediately after the main switch element Qa is turned on is shown. An operational result of the preceding timing s 10  is taken over, and the current Ia is flowing in the direction shown by the arrow. Also, the other main switch element Qb is off. At this time, the exciting current Im flows in the exciting inductance element Lm, in a direction opposite to that of the resonant current Ir. As it is bigger than the resonant current Ir, it flows as a discharge current in a direction discharging the charge of the resonant capacitor Cr. A current induced in the second coil L 2  by the ideal transformer Ti, based on the resonant current Ir of the primary side resonant circuit, is supplied to the load LD from the rectification diode D 1 . 
     In the operational condition in the timing s 2  shown in  FIG. 11 , the main switch element Qa continues to be on, and Qb continues to be off. However, the exciting current Im, which had been flowing in the direction opposite to that of the resonant current Ir, flows in the same direction as the resonant current Ir, and begins to charge the resonant capacitor Cr. Then, in the same way as in the preceding timing s 1 , the sinusoidal wave current I 1  is caused to flow in the secondary side rectification diode D 1  (refer to (F) of  FIG. 9 ) by the ideal transformer Ti, based on the resonant current Ir of the primary side resonant circuit, and power is supplied to the load LD. 
     In the timing s 3  shown in  FIG. 12 , the main switch element Qa continues to be on, and Qb continues to be off. However, a half cycle (Tr/2) of a resonance frequency Tr, determined by the size of the resonant inductor Lr and resonant capacitor Cr, has elapsed, and power has ceased to be supplied to the secondary side from the primary side of the ideal transformer Ti. However, as the operation frequency fop is lower than the resonance frequency fr 1 , and the main switch element Qa is in the on condition, the primary side resonant circuit resonates at the second resonance frequency fr 2 , and the resonant capacitor Cr is charged by the exciting current Im. As the cycle of the second resonance frequency fr 2  is extremely long in comparison with the cycle of the first resonance frequency fr 1 , the resonance waveform in the timing s 3  is approximately a straight line. 
     The timing s 4  shown in  FIG. 13  corresponds to a dead time when the two main switch elements Qa, Qb are both off. Herein, currents Ia, Ib flow, respectively, through a distributed capacity C_Qa of the main switch element Qa in the off condition, and a distributed capacity C_Qb of the other main switch element Qb in the off condition, in the directions shown by the arrows of  FIG. 13 , changing the voltage between the two ends of the two distributed capacities C_Qa and C_Qb. At this time, the resonant capacitor Cr, and the two distributed capacities C_Qa and C_Qb, form a resonant circuit of the resonant inductance Lr and exciting inductance element Lm, and carry out a resonance operation so as to charge the resonant capacitor Cr. However, no power is supplied to the secondary side from the primary side. As the capacitance of the resonant capacitor Cr is far bigger than the capacitance of the distributed capacities C_Qa, C_Qb, it is also possible, in the timing s 4 , to view the resonant capacitor Cr as a constant-voltage source. 
     In the timing s 5  shown in  FIG. 14 , the two main switch elements Qa, Qb continue to be in the off condition, but in the timing s 4  the voltage between the two ends of the distributed capacity C_Qb increases. When taking the side of the distributed capacity C_Qb connected to the resonant capacitor Cr in  FIG. 13  to be a reference potential, the absolute value increases at a negative voltage. As a result, in the main switch element Qb, a body diode D_Qb thereof attains a conduction state. At this time, by the resonant circuit resonating on the primary side, a current I 2  begins to flow from the primary side of the ideal transformer Ti, via the third coil L 3 , and a supply of power to the load LD begins. That is, as the exciting current Im flowing in the exciting inductance element Lm flows in a direction the reverse of that of the resonant current Ir as a bigger current than the resonant current Ir, the resonant capacitor Cr continues to be charged. As the current Ib is flowing in the body diode D_Qb (the orientation of the current is the direction shown by the arrow), and the other main switch element Qa is off, the terminal of the first coil L 1  of the transformer T on the side connected to the resonant capacitor Cr becomes a high voltage side, while the terminal on the opposite side is a low voltage side. 
     In the timing s 6  shown in  FIG. 15 , the main switch element Qb is in a condition immediately after being turned on, and the current Ib is flowing there, as before, in the direction shown by the arrow. Also, the other main switch element Qa is off. Consequently, the terminal of the first coil L 1  of the transformer Ton the side connected to the resonant capacitor Cr becomes the high voltage side (the terminal on the opposite side is the low voltage side), and power induced in the secondary side third coil L 3  from the primary side resonant circuit by the ideal transformer Ti is supplied to the load LD from the rectification diode D 2 . At this time, since the exciting current Im flows in the exciting inductance element Lm in a direction the reverse of that of the resonant current Ir, and is bigger than the resonant current Ir, the resonant capacitor Cr is charged by the current differential (Im-Ir). 
     In the timing s 7  shown in  FIG. 16 , the main switch element Qa continues to be off, and Qb continues to be in on. However, the exciting current Im becomes smaller than the resonant current Ir, and the exciting current Im, which had been flowing in the direction opposite to that of the resonant current Ir, flows in the same direction as the resonant current Ir (the direction shown by the arrow of  FIG. 16 ), and the charge accumulated in the resonant capacitor Cr begins to be discharged. Then, from the primary side resonant circuit, a sinusoidal wave current is caused to flow in the secondary side rectification diode D 2  (refer to (G) of  FIG. 9 ) by the ideal transformer Ti, and power is supplied to the load LD. 
     In the timing s 8  shown in  FIG. 17 , the main switch element Qa continues to be off, and Qb continues to be on. However, the half cycle resonance operation in the resonant inductor Lr and resonant capacitor Cr has finished, and power has ceased to be supplied to the secondary side from the primary side. Also, while the resonant capacitor Cr continues the discharge, the primary side resonant circuit carries out a resonance operation at the second resonance frequency fr 2 . 
     The timing s 9  shown in  FIG. 18  corresponds to a dead time when the two main switch elements Qa, Qb are both off. Herein, the currents Ia, Ib flow, respectively, through the distributed capacity C_Qa of the main switch element Qa, and the distributed capacity C_Qb of the other main switch element Qb, in the directions shown by the arrows of  FIG. 18 , changing the voltage between the two ends of the two distributed capacities C_Qa and C_Qb. At this time, the resonant capacitor Cr, and the two distributed capacities C_Qa, C_Qb configure a resonant circuit of the resonant inductance Lr and exciting inductance element Lm, and carry out a resonance operation so as to discharge the resonant capacitor Cr. However, no power is supplied to the secondary side from the primary side. As the capacitance of the resonant capacitor Cr is far bigger than the capacitance of the distributed capacities C_Qa, C_Qb, it is also possible, in the timing s 9 , to view the resonant capacitor Cr as a constant-voltage source. 
     In the timing s 10  shown in  FIG. 19 , the two main switch elements Qa, Qb continue to be in the off condition, but as a result of the voltage between the two ends of the distributed capacity C_Qa increasing in the timing s 9 , in the main switch element Qa, a body diode D_Qa thereof attains a conduction state. At this time, by the resonant circuit resonating on the primary side, a current I 1  begins to flow from the primary side of the ideal transformer Ti, via the second coil L 2 , and a supply of power to the load LD begins. Also, the exciting current Im flowing in the exciting inductance element Lm flows in a direction opposite to that of the resonant current Ir as a bigger current than the resonant current Ir, and the resonant capacitor Cr is further discharged. 
       FIG. 20  is an operation waveform diagram illustrating a current and voltage waveform of each portion in the second operational mode of the current resonant type converter. Herein, a description will be given of the second operational mode (Mode  2 ), in which the operation frequency fop is lower than the resonance frequency fr 1 , and the load LD is in the light load condition. 
     As shown in (A) and (B) of  FIG. 20 , a predetermined dead time is provided in the gate voltages Vga, Vgb of the main switch elements Qa, Qb. Also, as the load LD is in the light load condition, the currents Ia, Ib flowing in the main switch elements Qa, Qb decrease, and a fluctuation range of the inter-terminal voltage Vc of the resonant capacitor Cr shown in (E) of  FIG. 20  also decreases. Also, as the exciting current Im does not change so much from the time of the heavy load, the effect of the exciting current Im becomes relatively large, and there is more phase lead than at the time of the heavy load. Hereafter, a description will be given only of operational conditions differing from the circuit operations at the time of the heavy load shown in  FIG. 9 . 
       FIG. 21  shows an operational condition in a timing s 11  immediately after the main switch element Qa is turned on. This condition is a condition in which the main switch element Qa is on, and Qb is off. The resonant capacitor Cr forms a resonant circuit of the resonant inductance Lr and exciting inductance element Lm but, unlike in the timing s 1  (refer to  FIG. 10 ) in Mode  1  shown in FIG.  9 . Since the voltage between the two ends of the exciting inductance element Lm (the + side of  FIG. 21  is a high potential side) is low, power cannot immediately be supplied from the primary side to the secondary side, even though the main switch element Qa is turned on. 
     Subsequently, the resonant capacitor Cr discharges, and its inter-terminal voltage Vc decreases. Upon satisfying an Expression 3, at the end of the timing s 11 , the resonance operation in the primary side resonant circuit in the timing s 1  begins, and power is supplied from the primary side to the secondary side. Herein, Vf is a forward voltage of the secondary side rectification diodes D 1 , D 2 .
 
( Vi−Vc )* Lm /( Lr+Lm )= n *( V   0   +Vf )  (3)
 
       FIG. 22  illustrates an operational condition of a dead time timing s 12 , in which a half cycle (Top/2) of the switching operation has elapsed, and the two main switch elements Qa, Qb are both turned off. In this case, the body diode D_Qb of the main switch element Qb is conductive and, continuing from the preceding timing s 4 , the exciting current Im (to be precise, this is the resonant current of the resonant circuit configured of the resonant capacitor Cr, resonant inductor Lr, and exciting inductance element Lm) flows into the resonant capacitor Cr. However, as the voltage between the two ends of the exciting inductance element Lm (the + side of  FIG. 22  is a high potential side) is insufficient, unlike in the operational condition (timing s 5 ) shown in  FIG. 14 , it is not possible to supply power to the secondary side. 
       FIG. 23  illustrates the operational condition of the next timing s 13 . As shown here, the main switch element Qb is on, and the resonant capacitor Cr configures a resonant circuit of the resonant inductance Lr and exciting inductance element Lm. However, unlike the case of the timing s 6  of the first operational mode (Mode  1 ) shown in  FIG. 9 , as the voltage between the two ends of the exciting inductance element Lm (the + side of  FIG. 23  is a high potential side) is low, power cannot immediately be supplied from the primary side to the secondary side, even when the main switch element Qb is turned on. 
     Subsequently, the resonant capacitor Cr is charged. When its inter-terminal voltage Vc attains an operational condition (the condition of the timing s 6  of  FIG. 15 ) in which it satisfies an Expression 4, power is supplied from the primary side to the secondary side by a resonance operation of the resonant inductance Lr and resonant capacitor Cr.
 
 Vc*Lm /( Lr+Lm )= n *( V   0   +Vf )  (4)
 
     Subsequently, proceeding sequentially from the timing s 6  to the timings s 7 , s 8 , on arriving at the timing s 9  in which the two main switch elements Qa, Qb are both in the off condition, the currents Ia, Ib flow, respectively, through the distributed capacity C_Qa of the main switch element Qa, and the distributed capacity C_Qb of the other main switch element Qb, in the directions shown by the arrows of  FIG. 18 , changing the voltage between the two ends of the two distributed capacities C_Qa and C_Qb. At this time, the load LD is in the light condition. Therefore, in the timing s 14  in which the body diode D_Qa of the main switch element Qa is conductive, as shown in  FIG. 24 , the voltage between the two ends of the exciting inductance element Lm (the + side of  FIG. 24  is a high potential side) is insufficient. Unlike the operational condition (timing s 10 ) shown in  FIG. 19 , it is not possible to supply power to the secondary side. 
     Next, a description will be given for the third operational mode (Mode  3 ), in which the operation frequency fop is lower than the resonance frequency fr 1 , and the load LD is in an even smaller very light load (VLL) condition.  FIG. 25  is an operation waveform diagram illustrating a current and voltage waveform of each portion in the third operational mode of the current resonant type converter. In this case, the resonant current Ir further decreases, and the fluctuation range of the inter-terminal voltage Vc of the resonant capacitor Cr also further decreases. The voltage Vc of the resonant capacitor Cr is controlled by the exciting current Im (or by the resonant current of the resonant circuit configured of the resonant capacitor Cr, resonant inductor Lr, and exciting inductance element Lm), and there is even more phase lead. 
     In the timing s 2 , the kind of resonant current Ir and exciting current Im shown in  FIG. 11  flow, but the resonant current Ir is far smaller than the exciting current Im, and the inter-terminal voltage Vc of the resonant capacitor Cr is controlled by the exciting current Im. For this reason, the charge and discharge time by the resonant current Ir from the resonant inductor Lr becomes shorter. Consequently, in the third operational mode (Mode  3 ) in which the load becomes very light, a shift is made to the condition of the timing s 3  in a shorter time than in the first operational mode (the heavy load Mode  1 ) and second operational mode (the light load Mode  2 ). 
     In the same way, in the timing s 7 , the kind of resonant current Ir shown in  FIG. 16  is far smaller than the exciting current Im, and the inter-terminal voltage Vc of the resonant capacitor Cr is controlled by the exciting current Im. For this reason, the charge and discharge time by the resonant current Ir from the resonant inductor Lr becomes shorter, and in the operational mode with the very light load (Mode  3 ), a shift is made to the operational condition of the timing s 8  earlier than when there is a heavy load or a light load. 
     Furthermore, operation waveforms for the fourth to sixth operational modes (Modes  4  to  6 ), in which the operation frequency fop is higher than, or equal to, the resonance frequency fr 1 , are shown in  FIGS. 26 ,  29  and  30  respectively. 
       FIG. 26  is an operation waveform diagram illustrating a current and voltage waveform of each portion in the fourth operational mode of the current resonant type converter. In the fourth operational mode (Mode  4 ), the operation frequency fop is equal to or higher than the resonance frequency fr 1 , and the load LD is in a heavy load condition. 
       FIG. 27  illustrates an operational condition in a timing s 15  in  FIG. 26 , in which the two main switch elements Qa, Qb are turned off simultaneously. In the case of the fourth operational mode (Mode  4 ), even though the main switch element Qa is turned off in the timing s 15 , the body diode D_Qb of the main switch element Qb is conductive, and continues a discharge of energy accumulated in the resonant inductor Lr, but the + side potential of the exciting inductance element Lm is maintained. For this reason, the resonant current Ir flows in the ideal transformer Ti, and power is supplied from the primary side to the secondary side. In this case, as a voltage (n×(V 0 +V F )+Vc+V F ) is applied to the resonant inductor Lr, and it takes on a form approximating a constant voltage discharge, the current I 1 , which decreases approximately linearly, flows in the rectification diode D 1 . In the next timing s 5 , the resonant current Ir flows in the reverse direction, and the sinusoidal wave current I 2  begins to flow in the secondary side rectification diode D 2 . Consequently, in the fourth operational mode, the secondary side sinusoidal wave currents I 1 , I 2  flow consecutively. 
       FIG. 28 , in the same way, illustrates an operational condition in a timing s 16 , in which the two main switch elements Qa, Qb are turned off simultaneously. In this case, even though the main switch element Qb is turned off, the body diode D_Qa of the main switch element Qa is conductive. Also, due to the discharge of the resonant inductor Lr, the + side potential of the exciting inductance element Lm is maintained, and power is supplied to the secondary side. 
       FIG. 29  is an operation waveform diagram illustrating a current and voltage waveform of each portion in the fifth operational mode of the current resonant type converter. Herein, a description will be given of the fifth operational mode (Mode  5 ), in which the operation frequency fop is equal to or higher than the resonance frequency fr 1 , and the load LD is in a light load condition. 
     In each of the timings s 15 , s 16 , immediately after the two main switch elements Qa, Qb are turned off simultaneously, it is possible to supply power to the secondary side. However, as the load LD is in the light load condition, in each of the subsequent timings s 11 , s 12 , and s 13 , s 14 , in the same way as in the second operational mode (refer to  FIG. 20 ), a condition is such that no power is supplied to the secondary side. 
       FIG. 30  is an operation waveform diagram illustrating a current and voltage waveform of each portion in the sixth operational mode of the current resonant type converter. In Mode  6 , in which the operation frequency fop is equal to or higher than the resonance frequency fr 1 , and there is a very light load (VLL) condition, during a period in which the two main switch elements Qa, Qb are each turned on, the timings s 11 , s 3 , and s 13 , s 8  appear, in which no power is supplied to the secondary side. 
     Heretofore, a description has been given for the resonance operation in the six operational modes of the current resonant type converter shown in  FIG. 7 . Further consideration is made on a separately excited drive synchronous rectification in the current resonant type converter of  FIG. 6 , in which the secondary side rectification diodes D 1 , D 2  are replaced with the MOSFETs Qs 1 , Qs 2 , which have low resistances. 
     As synchronous rectification methods, there are a self excitation drive method and a separate excitation drive method. Regarding the separate excitation drive method, by building a logic circuit into a power source IC in order to emit a drive signal in the logic circuit, it becomes easy for a power source maker to realize a synchronous rectification function. Consequently, all IC manufacturers are devising various separate excitation drive methods (refer to U.S. Pat. No. 7,184,280: Patent Document 1, U.S. Patent Pub. App. No. 2008/0055942: Patent Document 2, U.S. Patent Pub. App. No. 2005/0122753: Patent Document 3, Japanese Unexamined Patent Application Publication No. JP-A-2005-198438: Patent Document 4, and Japanese Unexamined Patent Application Publication No. JP-A-2005-198375: Patent Document 5). 
     As to the separate excitation drive synchronous rectification, it seems to be sufficient that synchronous drive signals of the MOSFETs Qs 1 , Qs 2  are synchronized with the gate signals which conduct switching controls over the main switch elements Qa, Qb. However, in actual practice, if a reverse flow area is not detected in each operational mode and a conversion is not made to a drive signal synchronized with each one, the charge accumulated in the output capacitor C 0  is discharged, and a current flowing back to the transformer T side (a reverse current) occurs, thereby decreasing the efficiency. Furthermore, there emerges a danger of a circuit breakage due to the power flowing back to the primary side. 
     Herein, the reverse flow area in each operational mode is decided by the relationship between the operation frequency fop and resonance frequency fr 1  of the current resonant type converter, and the load LD. Of these, the operation frequency fop changes depending on circuit parameters and the load condition, but the resonance frequency fr 1  is determined by the size of the resonant capacitor Cr and resonant inductor Lr. Consequently, although a synchronous rectification which synchronizes the synchronous drive signal with a power switching signal is a simple method, in that case, a countermeasure is needed to eliminate the following five reverse flow areas. 
     That is, the switching power source device which turns the two main switch elements Qa, Qb on and off using the gate signals Vga, Vgb respectively, and supplies the secondary currents I 1 , I 2 , can prevent the secondary currents I 1 , I 2  from flowing back in the first operational mode (Mode  1 ) shown in  FIG. 31 , by reliably turning off the synchronous control MOSFETs Qs 1 , Qs 2  in the timings s 3 , s 8  of a latter half of the half cycle (Top/2) of the switching operation shown in  FIG. 9 . Consequently, in the event that the kind of gate signals Vga, Vgb shown in (A) and (B) in  FIG. 31  are output as they are to the synchronous control MOSFETs Qs 1 , Qs 2  as synchronous drive signals Vgs  1 , Vgs  2 , a reverse flow current flows in an area (Range A) of the timings s 3 , s 8 . 
     Also, in the case of the second operational mode (Mode  2 ), apart from Range A in which the reverse current occurs, there is also a danger of the reverse current occurring in an area Range B shown in  FIG. 32  (corresponding to the timings s 11  and s 13  shown in  FIG. 20 ). 
     In the same way, in the case of the third operational mode (Mode  3 ), as shown in  FIG. 33 , as well as the reverse current occurring in Range A and Range B, the reverse current also occurs in Range C (this is within a half cycle of a resonance cycle Tr, but corresponds to an area in which the resonance is finished). 
     In the case of the fourth operational mode (Mode  4 ) shown in  FIG. 34 , there is no danger of the reverse current occurring, because the secondary currents I 1 , I 2  are consecutive. 
     In the fifth operational mode (Mode  5 ) shown in  FIG. 35 , the reverse current occurs in an area Range D (corresponding to the timings s 11 , s 13  shown in  FIG. 29 ). 
     In the case of the sixth operational mode (Mode  6 ), as shown in  FIG. 36 , as well as the reverse current occurring in Range D, the reverse current also occurs in an area Range E (corresponding to the timings s 3  and s 8  shown in  FIG. 30 ). Consequently, in the event of applying signals synchronized with the gate signals Vga, Vgb (the same signals) as the synchronous drive signals Vgs 1 , Vgs 2  to the synchronous control MOSFETs Qs 1 , Qs 2 , as the reverse current occurs in each of the operational modes  1  to  3 , and  5  and  6 , it has been necessary to form individual Vgs  1  and Vgs  2  signal waveforms in the areas (Ranges A to E) corresponding thereto. 
     Therefore, with the heretofore known switching power source devices, there is provided a constant width pulse (CWP) generation circuit which outputs a CWP signal with a pulse width slightly narrower than an on period of the gate signals Vga, Vgb, forming the waveforms of the synchronous drive signals Vgs  1 , Vgs 2 . That is, when the operation frequency fop is the same as, or higher than, the resonance frequency fr 1 , the synchronous drive signals Vgs  1 , Vgs 2  are synchronized with the gate signals Vga, Vgb, and when the operation frequency fop is lower than the resonance frequency fr 1 , the synchronous drive signals Vgs  1 , Vgs 2  are synchronized with the constant width pulse signal CWP, causing it to finish (for example, refer to Patent Document 1). Because of this, even in the case of replacing the secondary side rectification diodes D 1 , D 2  with the MOSFETs Qs 1 , Qs 2 , which have low on resistance, it is possible to prevent the reverse flow current from the secondary side. 
     However, with the invention described in Patent Document 1, as a timing of a start-up of the synchronous drive signals Vgs  1 , Vgs 2  is always synchronized with the gate signals Vga, Vgb, it is difficult to prevent the reverse current immediately before the secondary current begins to flow, as in the reverse current area (Range B) in the second operational mode (Mode  2 ). Also, in the cases of Modes  4  to  6 , in which the operation frequency fop is the same as, or higher than, the resonance frequency fr 1 , in the event the synchronous drive signals Vgs 1 , Vgs 2  are synchronized with the gate signals Vga, Vgb, it is not possible to prevent the reverse current in the light load (LL) condition or very light load (VLL) condition. 
     As a different switching power supply device, a method for comprising the kind of synchronous rectification MOSFET control circuit shown in  FIG. 37(A)  is known (for example, Patent Document 2). Also, an operation waveform of each portion thereof is shown in  FIG. 37(B) . This is a method which compares a drain-to-source voltage (Vds (on)) of a synchronous rectification switch element (MOSFET) with a reference voltage REF in a comparator  510 , detects that the synchronous rectification MOSFET, or a body diode thereof, is conductive and, provides a signal to the synchronous rectification MOSFET causing the synchronous rectification MOSFET to be turned on only for a period that the conductivity is detected and a gate signal Vgp is high (H). That is, the method generates a comparison signal Vdsc, which is an output of the comparator  510 , in an AND circuit  430 , and an AND signal of the gate signal Vgp of the primary side main switch elements Qa, Qb, and outputs them as synchronous drive signals Vgs (that is, Vgs 1  and Vgs 2 ), of which the waveform has been formed, to the MOSFETs Qs 1  and Qs 2 , which are the switch elements. 
     Generally, the drain-to-source voltage Vds of the MOSFET, in a condition in which the MOSFET is turned off and a current is flowing in the body diode, becomes a body diode forward drop voltage V F  (to be precise, taking a source potential as a reference potential, it is −V F ). Meanwhile, in a condition in which the MOSFET is turned on, the drain-to-source voltage is a product of the on resistance of the MOSFET and the current flowing, and a value (an absolute value) thereof is normally smaller than V F . The reference voltage REF firstly, detecting that a current is flowing in the body diode, allows the MOSFET to be turned on, and subsequently, the MOSFET being turned on, in order that it is possible to cause the MOSFET to continue to be turned on even when the drain-to-source voltage Vds is small, the absolute value of the reference voltage REF is made considerably small (actually, taking noise and the like into consideration, it is necessary to make the absolute value large enough to be able to detect without error that the MOSFET, or the body diode thereof, is conductive). 
     However, as shown in  FIG. 37(B) , when the secondary current Is decreases to zero, how small the value of the reference voltage REF is, at some point the product of the on resistance of the MOSFET and the current flowing will become smaller. On this happening, a condition is such that the comparison signal Vdsc inverts, the MOSFET is turned off, and a current flows in the body diode, whereon the drain-to-source voltage Vds becomes −V F . Because of this, the comparison signal Vdsc inverts again, and the MOSFET is turned on again, as a result of which the comparison signal Vdsc further inverts. Subsequently, as shown in the error area of  FIG. 37(B) , the turning on and off of the MOSFET is repeated at a high frequency until the secondary current Is is definitely zero. This resonance phenomenon becomes more noticeable as the load becomes lighter, and the secondary current Is decreases. As such, as a high frequency resonance is repeated every time the secondary current Is decreases to zero, the invention described in Patent Document 1 is a method having a problem from the point of view of noise and power conversion efficiency. 
     As an invention which takes into consideration a conducting voltage of the internal diode, and sets a turn on threshold value (V TH2 ), there is Patent Document 3. Herein, as a turn on timing of the synchronous drive signal is decided only by the conducting voltage of the internal diode, there is a problem in that a malfunction is liable to occur in the dead time set in the primary side gate signals Vga, Vgb. Also, as a threshold value (V TH1 ) which determines a turn off timing is a minute voltage value of around −20 mV, and what is more a negative value, there is a problem in that it is easily effected by noise, and the timing of the off operation is unstable. 
     Also, with a different switching power source device, the primary side resonant current is detected with a current transformer, the exciting current is detected with a secondary side auxiliary coil, and a resonant current detection signal is compared with an exciting current detection signal. A synchronous rectification signal is generated based on a signal detecting whether or not a comparison result signal, a power switching signal, and the resonant current detection signal exceed 0 A (for example, Patent Document 4). 
     With the technology of Patent Document 4, it is possible to solve the reverse flow problem in each non-consecutive mode but, as the on timing of the synchronous rectification MOSFETS is delayed in the operational modes (Modes  1  and  4 ) with the heavy load condition, the power efficiency decreases. Moreover, as the current transformer and auxiliary coil are used in the detection circuit, a circuit configuration becomes complex, and so on, it is difficult to design to an appropriate adjusted value, and it is not desirable from a point of view of cost either. 
     Furthermore, as a synchronous rectification circuit which may prevent the current from flowing in the reverse direction, and a power converter which attempts a reduction in power conversion loss, there is the invention described in Patent Document 5. This invention compares a synchronous rectification transistor source-to-drain voltage in a comparator circuit and, when detecting a reverse direction current, attempts to prevent it using a switching unit. Herein, although a timing in which the synchronous rectification transistor is turned off is decided, there is no mention of a timing in which it is turned on. Consequently, the invention is not effective as a measure for preventing the reverse flow of the current (Ranges B and D) in the second operational mode (Mode  2 ), third operational mode (Mode  3 ), fifth operational mode (Mode  5 ), and sixth operational mode (Mode  6 ). 
     As such, with the heretofore known switching power source devices, there is no drive circuit which reliably prevents the reverse flow of the secondary current to the primary side in all six of the operational modes (refer to  FIG. 8 ). In particular, there has been a need to execute a simple synchronous rectification which prevents the current reverse flow in the light load condition, and also prevents a malfunction by stably detecting the synchronous rectification MOSFET drain-to-source voltage (Vds). 
     The invention has been made in view of such problems. Therefore, an object of the present invention is to provide a switching power source device, a switching power source control circuit, and a switching power source device control method which can prevent a reverse flow of a current in any operational mode, and further realize a stable synchronous rectification function. 
     Further objects and advantages of the invention will be apparent from the following description of the invention. 
     SUMMARY OF THE INVENTION 
     In the invention, in order to solve the heretofore described problem, there is provided a switching power source device wherein an input direct voltage is applied to a series resonant circuit, the device generates a predetermined output voltage via a transformer and supplies power to a load. 
     In the switching power source device, the series resonant circuit includes a current resonant inductor and a current resonant capacitor. Also, a plurality of main switch elements or main switch element groups, comprising, for example, MOSFETs, switch a current path of the series resonant circuit by alternately turning on and off. A transformer induces a current from the series resonant circuit to a secondary side by on-off controlling the main switch elements or main switch element groups on a primary side. A plurality of synchronous rectification switch elements, in which internal diodes are connected in parallel, are each turned on and off in response to either the plurality of main switch elements or main switch element groups, rectifying the secondary current of the transformer. A maximum on width control circuit, in synchronization with a timing in which the main switch elements or main switch element groups are turned on, indicates a start of a maximum on width to the synchronous rectification switch element and, after a predetermined time, indicates a completion of the maximum on width. A synchronous control circuit controls an on period of the synchronous rectification switch element in such a way as to turn on the synchronous rectification switch element in synchronization with a timing in which the maximum on width control circuit indicates the start of the maximum on width, or a timing of a conduction of the internal diodes detected by an inter-terminal voltage signal of the synchronous rectification switch element, whichever timing is later, and turn off the synchronous rectification switch element in synchronization with a timing in which the main switch elements or main switch element groups are turned off, or a timing in which the maximum on width control circuit indicates the completion of the maximum on width, whichever timing is earlier. 
     According to the switching power source device, the switching power source control circuit, and the switching power source device control method of the invention, since the on period of the synchronous rectification switch element is controlled by utilizing a gate on-off signal and maximum on width signal applied to a gate of the main switch elements or main switch element groups, it is possible to eliminate all noise except at a time when the main switch elements or main switch element groups are turned on. 
     Also, the conduction voltage of the diodes connected in parallel in the synchronous rectification switch element is detected from the synchronous rectification switch element inter-terminal voltage level, and is used only in the synchronous rectification switch element turn on timing control, and the maximum on width signal is effectively applied. Therefore, it is possible to provide a switching power source device control method, a switching power source control circuit, and a switching power source device which realize a synchronous rectification function of a current resonant type converter which is not affected by inter-terminal voltage level detection noise, has no malfunction, and in which no reverse flow occurs. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a circuit diagram illustrating an overall configuration of a switching power source device according to an embodiment; 
         FIG. 2  is a circuit diagram illustrating a maximum on width control circuit and a synchronous control circuit of the switching power source device; 
         FIG. 3  is a timing diagram illustrating a secondary side current control operation by the synchronous control circuit of  FIG. 2 ; 
         FIG. 4  is a circuit diagram illustrating an overall configuration of a full bridge type of switching power source device; 
         FIG. 5  is a circuit diagram illustrating one example of a conventional current resonant type converter; 
         FIG. 6  is a diagram illustrating the current resonant type converter in which a rectification diode of  FIG. 5  is replaced with a MOSFET; 
         FIG. 7  is a circuit diagram illustrating one example of the current resonant type converter of a secondary side diode rectification; 
         FIG. 8  is a diagram illustrating a relationship between an operation frequency fs and a resonance frequency fr of six operational modes, and a load condition; 
         FIG. 9  is an operation waveform diagram illustrating a current and voltage waveform of each portion in a first operational mode of the current resonant type converter; 
         FIG. 10  is a circuit diagram illustrating an operational condition in a timing s 1  of the first operational mode; 
         FIG. 11  is a circuit diagram illustrating an operational condition in a timing s 2  of the first operational mode; 
         FIG. 12  is a circuit diagram illustrating an operational condition in a timing s 3  of the first operational mode; 
         FIG. 13  is a circuit diagram illustrating an operational condition in a timing s 4  of the first operational mode; 
         FIG. 14  is a circuit diagram illustrating an operational condition in a timing s 5  of the first operational mode; 
         FIG. 15  is a circuit diagram illustrating an operational condition in a timing s 6  of the first operational mode; 
         FIG. 16  is a circuit diagram illustrating an operational condition in a timing s 7  of the first operational mode; 
         FIG. 17  is a circuit diagram illustrating an operational condition in a timing s 8  of the first operational mode; 
         FIG. 18  is a circuit diagram illustrating an operational condition in a timing s 9  of the first operational mode; 
         FIG. 19  is a circuit diagram illustrating an operational condition in a timing s 10  of the first operational mode; 
         FIG. 20  is an operation waveform diagram illustrating a current and voltage waveform of each portion in a second operational mode of the current resonant type converter; 
         FIG. 21  is a circuit diagram illustrating an operational condition in a timing s 11  of the second operational mode; 
         FIG. 22  is a circuit diagram illustrating an operational condition in a timing s 12  of the second operational mode; 
         FIG. 23  is a circuit diagram illustrating an Operational condition in a timing s 13  of the second operational mode; 
         FIG. 24  is a circuit diagram illustrating an operational condition in a timing s 14  of the second operational mode; 
         FIG. 25  is an operation waveform diagram illustrating a current and voltage waveform of each portion in a third operational mode of the current resonant type converter; 
         FIG. 26  is an operation waveform diagram illustrating a current and voltage waveform of each portion in a fourth operational mode of the current resonant type converter; 
         FIG. 27  is a circuit diagram illustrating an operational condition in a timing s 15  of the fourth operational mode; 
         FIG. 28  is a circuit diagram illustrating an operational condition in a timing s 16  of the fourth operational mode; 
         FIG. 29  is an operation waveform diagram illustrating a current and voltage waveform of each portion in a fifth operational mode of the current resonant type converter; 
         FIG. 30  is an operation waveform diagram illustrating a current and voltage waveform of each portion in a sixth operational mode of the current resonant type converter; 
         FIG. 31  is a diagram illustrating a secondary side current waveform induced via a transformer in the first operational mode; 
         FIG. 32  is a diagram illustrating a secondary side current waveform induced via a transformer in the second operational mode; 
         FIG. 33  is a diagram illustrating a secondary side current waveform induced via a transformer in the third operational mode; 
         FIG. 34  is a diagram illustrating a secondary side current waveform induced via a transformer in the fourth operational mode; 
         FIG. 35  is a diagram illustrating a secondary side current waveform induced via a transformer in the fifth operational mode; 
         FIG. 36  is a diagram illustrating a secondary side current waveform induced via a transformer in the sixth operational mode; and 
         FIGS. 37(A) and 37(B)  being diagrams for illustrating problem areas of heretofore known technology,  FIG. 37(A)  is a diagram illustrating a control circuit of a synchronous rectification MOSFET, and  FIG. 37(B)  is an operation waveform diagram of each portion of the control circuit. 
     
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     Hereafter, referring to the drawings, a description will be given of embodiments of the invention.  FIG. 1  is a circuit diagram illustrating an overall configuration of a switching power source device according to the embodiment. 
     The switching power source device is configured in such a way that an input direct voltage Vi is applied to a series resonant circuit which includes a resonant inductor Lr and a resonant capacitor Cr, generating a predetermined output voltage V 0  on a load LD via a transformer T. It is also acceptable, not providing any inductance other than the transformer T, to have the current resonant inductor Lr as leakage inductance. Alternatively, it is also acceptable to provide external inductance other than the transformer T, and configure the current resonant inductor Lr as a combination of the external inductance and leakage inductance. MOSFETs Qa and Qb are main switch elements which, by switching alternately on a primary side of the transformer T, switch a current path to the series resonant circuit. One extremity of the resonant inductor Lr is connected to one extremity of a first coil L 1  of the transformer T, while the other extremity of the first coil L 1  is connected to one extremity of the resonant capacitor Cr. 
     A second coil L 2  and a third coil L 3  being mutually connected in series on a secondary side of the transformer T, their connection point is connected to one extremity of an output capacitor C 0  and the load LD. Also, the other extremities of the second coil L 2  and third coil L 3  are connected to the other extremity of the ground side of the output capacitor C 0  and load LD via MOSFETs Qs 1 , Qs 2  respectively. The MOSFETs Qs 1 , Qs 2  being synchronous rectification switch elements which supply the predetermined output voltage V 0  to the load LD by turning on and off secondary currents  11  and  12 , induced from the transformer T, in response to operations of the MOSFETs Qa, Qb, the kinds of internal diode (body diodes, or body diodes and external diodes) Ds to be described hereafter are connected in parallel to each of them. 
     The output voltage V 0  supplied to the load LD is returned to a drive circuit  3  of the MOSFETs Qa, Qb via an error amplifier  1  and a voltage control oscillation circuit (VCO)  2 . At this time, gate signals Vga, Vgb, which alternately turn on and off in a predetermined timing, are generated in the drive circuit  3  and, by controlling in such a way as to send currents Ia, Ib of the MOSFETs Qa, Qb in the directions of the arrows in a predetermined timing, control a current and voltage flowing in the first coil L 1  of the transformer T, and control the secondary side output voltage V 0  at a constant voltage. The VCO  2  functions in such a way that, when determining that due to an output of the error amplifier  1  the output voltage V 0  is higher than a set voltage, or is a light load, it increases an output frequency thereof, while when determining that the output voltage V 0  is lower than the set voltage, or is a heavy load, it reduces the output frequency thereof. 
     In maximum on width control circuits  41 ,  42 , in synchronization with an on timing of the MOSFETs Qa, Qb, a maximum on width signal Tmot which is a maximum on width period H (a period other than this is low (L)), for indicating a maximum on width of a predetermined time for which the MOSFETs Qs 1 , Qs 2  are mandatorily turned off on exceeding the period, or a maximum on width completion signal Tmot  2  which is a signal indicating a completion of the maximum on width, is generated and output to the synchronous rectification MOSFETs Qs 1 , Qs 2 . A start of the maximum on width, being indicated by the gate signals Vga, Vgb to the MOSFETs Qa, Qb, is the same as a timing in which the gate signals Vga, Vgb become H, and the MOSFETs Qa, Qb are turned on. Then, in synchronous control circuits  51 ,  52 , which actually generate synchronous drive signals Vgs  1 , Vgs  2  of the synchronous rectification MOSFETs Qs 1 , Qs 2 , a timing of the MOSFETs Qs 1 , Qs 2  being turned on is decided synchronized with a timing indicating the start of the maximum on width (that is, a timing of the maximum on width signal Tmot changing from L to H), or a timing of a conduction of the internal diodes Ds detected by drain-to-source voltages Vds 1  and Vds 2  of the MOSFETs Qs 1 , Qs 2 , whichever timing is later, and furthermore, a timing of the MOSFETs Qs 1 , Qs 2  being turned off is decided synchronized with an off timing of the MOSFETs Qa, Qb, or a timing of indicating the completion of the maximum on width, whichever timing is earlier. 
     Next, a description will be given of a control circuit (a switching power source control circuit) of the switching power source device shown in  FIG. 1 . 
       FIG. 2  is a circuit diagram illustrating a maximum on width control circuit and synchronous control circuit of the switching power source device. Herein, the synchronous control circuit  5  is illustrated representing either one of the synchronous control circuits  51 ,  52 . Also, with regard to the maximum on width control circuits  41  and  42 , by configuring them of a one-shot multivibrator which adjusts its pulse width by means of an external part connected to an MOT terminal, the maximum on width control circuits  41 ,  42  can realize an embodiment which outputs the maximum on width signal Tmot, and this is taken as a first embodiment. However, as the one-shot multivibrator itself is commonly known, any further description relating to the first embodiment will be omitted. Herein, a description will be given of a second embodiment realized by the maximum on width control circuit  4  which operates in conformity with the first embodiment. The maximum on width control circuit  4  illustrated in  FIG. 2  is illustrated representing either one of the maximum on width control circuits  41  or  42 . Also, in the case of realizing the maximum on width control circuits  41 ,  42  with a one-shot multivibrator too, it is possible to use the same synchronous control circuit  5 . Also, the internal diodes Ds are connected in parallel to a drain-to-source of a synchronous rectification MOSFET Qs, which controls a secondary current Is of either one of the second coil L 2  or third coil L 3  (described as Ls in  FIG. 2 ) on the secondary side of the transformer T. 
     The maximum on width control circuit  4  includes a gate signal input terminal  4   a , to which either one of the gate signals Vga or Vgb (hereafter described simply as a gate signal Vgp) is supplied from the drive circuit  3  shown in  FIG. 1 , and an MOT terminal  4   b , to which is connected an external part for adjusting the size of the maximum on width (corresponding to the pulse width of the maximum on width signal Tmot, which is the output of the one-shot multivibrator when using the one-shot multivibrator). The maximum on width control circuit  4  is a circuit which, rather than outputting the maximum on width signal Tmot itself, outputs the maximum on width completion signal Tmot  2  indicating the timing of the completion of the maximum on width. The maximum on width control circuit  4  is configured of an inverter  43  connected to the gate signal input terminal  4   a , a constant current source  44  connected to a power source voltage VDD, a capacitor C 1 , of which one extremity is connected to the constant current source  44  and the other extremity is grounded, a switch S 1 , which is on-off controlled by the inverter  43  in such a way as to control a charge and discharge of the capacitor C 1 , and a comparator  45  which compares a charging voltage to the capacitor C 1  with a reference voltage REF  1 . 
     In the maximum on width control circuit  4 , on the switch S 1  being turned off by the gate signal Vgp of the gate signal input terminal  4   a , the current from the constant current source  44  starts to charge the capacitor C 1 . Then, in a timing in which the voltage of an inverting input terminal (−) of the comparator  45  exceeds the reference voltage REF  1  to a non-inverting input terminal (+), the maximum on width completion signal Tmot  2 , which is the output of the comparator  45 , reverses from H (high) to L (low), and communicates the completion of the maximum on width to the synchronous control circuit  5 . On the maximum on width completion signal Tmot  2  becoming L, the output of a negative AND (NAND) circuit  54  of the synchronous control circuit  5  becomes H, resetting a flip-flop circuit  55 . Subsequently, on the switch S 1  being turned on by the gate signal Vgp, the capacitor C 1  is discharged, and the maximum on width completion signal Tmot  2  becomes H. The start of the maximum on width signal Tmot being in the timing in which the gate signal Vgp becomes H, as heretofore described, the gate signal Vgp is input into the NAND circuit  54  connected to a reset terminal of the flip-flop circuit  55 , by which means the maximum on width control circuit  4  of the second embodiment can indicate the start of the maximum on width (the completion of the resetting of the flip-flop circuit  55 ). Consequently, in the second embodiment too, it is possible to indicate the same timing for the start and completion of the maximum on width as with the maximum on width signal Tmot when using the one-shot multivibrator of the first embodiment, because of which it is possible to set a maximum on width having an optimum on width. 
     A resistor, a capacitor, or the like is connected to the MOT terminal  4   b  for adjusting the output timing of the maximum on width completion signal Tmot  2 , output from the comparator  45 , in accordance with a resonance frequency fr 1 . In the case of connecting a resistor, the current from the constant current source  44  is split by the resistor, and it is possible to delay the output timing of the maximum on width completion signal Tmot  2  in accordance with a resistance value of the resistor. Also, in the case of connecting a capacitor, it is equivalent to a capacity value of the capacitor C 1  increasing, because of which it is possible to delay the output timing of the maximum on width completion signal Tmot  2 . 
     Herein, as a logical product (a negative thereof) of the gate signal Vgp and maximum on width completion signal Tmot  2  is taken in the NAND circuit  54 , the flip-flop circuit  55  is reset in either the timing in which the maximum on width completion signal Tmot  2  becomes L, or the timing in which the on period of the gate signal Vgp is completed, and the gate signal Vgp becomes L, whichever is earlier. 
     The synchronous control circuit  5  is a circuit which, including a signal input terminal  5   a  to which is supplied a very light load condition signal VLL which becomes H when a very light load is detected by a circuit (not shown), and furthermore, to which are supplied the drain-to-source voltage Vds from the synchronous rectification MOSFET Qs, the maximum on width signal Tmot or the maximum on width completion signal Tmot  2  output from the maximum on width control circuit  4 , and the gate signal Vgp which on-off controls the MOSFETs Qa, Qb, controls an on period of the MOSFET Qs by outputting the synchronous drive signal Vgs. The synchronous control circuit  5  is configured of a comparator  53  which, based on a reference voltage REF  2 , outputs a level detection signal Vdsc from the drain-to-source voltage Vds, the NAND circuit  54 , which computes NAND signals of the maximum on width signal Tmot and gate signal Vgp, the flip-flop circuit  55 , in which the level detection signal Vdsc of the comparator  53  and the NAND signals of the NAND circuit  54  are supplied to a set terminal S and a reset terminal R respectively, and an AND circuit  57 , into which are input a Q output signal of the flip-flop circuit  55 , and a signal which is the very light load condition signal VLL to the signal input terminal  5   a  inverted by an inverter  56 . 
     In the synchronous control circuit  5 , an inverting input terminal (−) of the comparator  53 , as well as being grounded via a Zener diode ZD, is connected to the power source voltage VDD via a resistor R 1 , and furthermore, is connected to a drain terminal of the synchronous rectification MOSFET Qs via a resistor R 2 . The reference voltage REF  2  of the comparator  53  is applied to a non-inverting input terminal (+). 
     Now, taking the power source voltage VDD to be A, and a threshold value voltage (Vds_th) which is the MOSFET Qs drain-to-source voltage Vds when the level detection signal Vdsc, which is the output of the comparator  53 , inverts to be X, as the two inputs of the comparator  53  are equal when Vds=X, the following expression holds true. The above mentioned threshold value voltage (Vds_th) is different from a threshold value relating to an on-off of the MOSFET Qs.
 
( A−X )*( R 2/( R 1+ R 2)+ X =REF2
 
∴( A−X )+ X *(1+ R 1 /R 2)=REF2*(1+ R 1 /R 2)
 
∴ X *( R 1/ R 2)=(1+ R 1 /R 2)*REF2 −A  
 
     Consequently, the MOSFET Qs drain-to-source voltage threshold value voltage (Vds_th) when the level detection signal Vdsc inverts is, 
     
       
         
           
             
               
                 
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     When the drain-to-source voltage Vds exceeds (Vds_th) (strictly speaking, when both are negative values, and an absolute value of Vds is greater than an absolute value of (Vds_th)), the level detection signal Vdsc, which is the output of the comparator  53 , inverts from L to H. Incidentally, in a condition in which no current is flowing in the drain-to-source, the value of the drain-to-source voltage Vds is positive, and the level detection signal Vdsc is L. While the level detection signal Vdsc of the drain-to-source voltage Vds is supplied as a set signal of the flip-flop circuit  55 , an output signal of the NAND circuit  54 , into which are input the gate signal Vgp, and the maximum on width signal Tmot or maximum on width completion signal Tmot  2 , is taken to be a reset signal of the flip-flop circuit  55 . Also, the Q output signal of the flip-flop circuit  55  being supplied to the AND circuit  57  along with the inverted very light load condition signal VLL output from the inverter  56 , the output signal is taken to be the gate drive signal Vgs of the synchronous rectification MOSFET Qs. The flip-flop circuit  55  is a reset prioritizing circuit. That is, when the gate signal Vgp is in the L condition before indicating the start of the maximum on width, it is not possible to set the flip-flop circuit  55 , even in the event that the drain-to-source voltage Vds exceeds (Vds_th). Because of this, it is possible to prevent a malfunction which may occur in a dead time set in the primary side gate signals Vga, Vgb. 
     Next, a description will be given of a control method of the switching power source device. 
       FIG. 3  is a timing diagram illustrating a secondary side current control operation by the synchronous control circuit of  FIG. 2 . Herein, a description is given of a case in which the maximum on width control circuits  41 ,  42  are realized with a one-shot multivibrator. In (A) of  FIG. 3 , the gate signal Vgp from the drive circuit  3  shown in  FIG. 1  is shown for six operational modes, Modes  1  to  6 , lined along the same time axis. (B) of  FIG. 3  shows the secondary current Is flowing in the synchronous rectification MOSFET Qs compared with an absolute value (|Vds_th|) of the MOSFET Qs drain-to-source voltage when the level detection signal Vdsc inverts converted into a current by dividing by an on resistance Ron of the MOSFET Qs. Also, the threshold value voltage Vds_th can be calculated using the heretofore described Expression 5. Alternatively, it is also possible to decide the voltage Vds_th first, and adjust other parameters using the Expression 5. 
     In (C) of  FIG. 3 , a waveform of the level detection signal Vdcs is shown. The |Vds_th|/Ron shown in (B) of  FIG. 3  shows a level of the secondary current Is for which the level detection signal Vdsc output from the comparator  53  is H. Also, (D) of  FIG. 3  is the maximum on width signal Tmot, (E) of  FIG. 3  the very light load condition signal VLL, and (F) of  FIG. 3  the synchronous drive signal Vgs output from the synchronous control circuit  5 . 
     In the first operational mode (Mode  1 ), the synchronous drive signal Vgs turns off the MOSFET Qs in synchronization with the timing in which the gate signal Vgp is turned off, or the timing in which the maximum on width signal Tmot indicates off, whichever timing is earlier. For this reason, it is possible to reliably prevent a reverse flow in a Range A which has heretofore occurred in a latter half of a half cycle (Top/2) of a switching operation. 
     In the second and fifth operational modes (Modes  2  and  5 ), the synchronous drive signal Vgs turning on the MOSFET Qs in synchronization with the timing in which the maximum on width signal Tmot indicates on, or the timing in which the conduction of the internal diodes Ds detected by the level detection signal Vdsc of the drain-to-source voltage Vds, whichever timing is later, it is also possible to reliably prevent a reverse flow in a Range B ( FIG. 32 ) or a Range D ( FIG. 35 ) which has heretofore occurred in a first half of a half cycle (Top/2) of the switching operation. Also, once the MOSFET Qs has been turned on, a fluctuation of the drain-to-source voltage Vds is ignored. Consequently, there is no occurrence of the phenomenon seen in the configuration of Patent Document 2 wherein a high frequency oscillation is invariably repeated every time the secondary current Is decreases to zero. 
     Also, as the very light load condition of the load LD is detected by the very light load condition signal VLL, the synchronous drive signal Vgs is not output from the synchronous control circuit  5  in the third and sixth operational modes (Modes  3  and  6 ). For this reason, the MOSFET Qs not being turned on in the very light load condition, it is also possible to reliably prevent a reverse flow which has heretofore occurred in a Range C ( FIG. 33 ) or a Range E ( FIG. 36 ). The very light load condition signal VLL has been defined as a case in which the load LD actually connected is 20% or less of a rated load (maximum load) of the switching power source device, but the setting of this ratio can be changed as appropriate. 
     As heretofore described, as it is possible to supply the synchronous drive signal Vgs to the synchronous rectification MOSFET Qs with the kind of timing shown in  FIG. 3 , a stable synchronous rectification function is realized, with no reverse current occurring, in the case of any of the first to sixth operational modes. 
     In the heretofore described embodiments, a description is given of a half bridge type of switching power source device, but the invention is also applicable to a full bridge type of switching power source device or switching power source control circuit, and to a control method of the full bridge type of switching power source device. 
       FIG. 4  is a circuit diagram illustrating an overall configuration of the full bridge type of switching power source device. 
     In the full bridge type of switching power source device shown in  FIG. 4 , the gate signals Vga, Vgb, which alternately turn on and off in a predetermined timing, are generated in the drive circuit  3 , and a first main switch element group of MOSFETs Qa 1 , Qa 2  and second main switch element group of MOSFETs Qb 1 , Qb 2  are alternately switched by the gate signals Vga, Vgb on the primary side of the transformer T. A current Ia flowing in the direction of the arrow in the timing in which the first main switch element group of MOSFETs Qa 1 , Qa 2  is turned on, and a current Ib flowing in the direction of the arrow in the timing in which the second main switch element group of MOSFETs Qb 1 , Qb 2  is turned on, in each case the input direct voltage Vi is applied to the series resonant circuit which includes the resonant inductor Lr and resonant capacitor Cr. 
     On the secondary side of the transformer T, the MOSFETs Qs 1 , Qs 2 , acting as synchronous rectification switch elements, supply the predetermined output voltage V 0  to the load LD by turning on and off the secondary currents  11 ,  12 , induced from the transformer T, in response to operations of the first main switch element group of MOSFETs Qa 1 , Qa 2 , or of the second main switch element group of MOSFETs Qb 1 , Qb 2 . 
     In the maximum on width control circuits  41 ,  42 , in synchronization with an on timing of the first main switch element group or second main switch element group of MOSFETs Qa 1 , Qa 2 , Qb 1 , and Qb 2 , the maximum on width signal Tmot which is the maximum on width period H (a period other than this is low (L)), for indicating the maximum on width of the predetermined time for which the MOSFETs Qs 1 , Qs 2  are mandatorily turned off on exceeding the period, or the maximum on width completion signal Tmot  2  which is a signal indicating the completion of the maximum on width, is generated and output to the synchronous rectification MOSFETs Qs 1 , Qs 2 . Herein too, the start of the maximum on width, being indicated by the gate signals Vga, Vgb to the MOSFETs Qa 1 , Qa 2 , Qb 1 , and Qb 2 , is the same as the timing in which the gate signals Vga, Vgb become H, and the MOSFETs Qa 1 , Qa 2 , Qb 1 , and Qb 2  are turned on. Then, in the synchronous control circuits  51 ,  52 , which actually generate the synchronous drive signals Vgs  1 , Vgs  2  of the synchronous rectification MOSFETs Qs 1 , Qs 2 , the timing of the MOSFETs Qs 1 , Qs 2  being turned on is decided synchronized with the timing indicating the start of the maximum on width (that is, the timing of the maximum on width signal Tmot changing from L to H), or a timing of the conduction of the internal diodes Ds detected by the drain-to-source voltages Vds 1 , Vds 2  of the MOSFETs Qs 1 , Qs 2 , whichever timing is later, and furthermore, the timing of the MOSFETs Qs 1 , Qs 2  being turned off is decided synchronized with the off timing of the MOSFETs Qa 1 , Qa 2 , Qb 1 , and Qb 2 , or the timing of indicating the completion of the maximum on width, whichever timing is earlier. 
     The switching power source control circuit and the control method thereof being the same as in the case of the half bridge type of switching power source device, their descriptions will be omitted. 
     The disclosure of Japanese Patent Application No. 2009-132333 filed on Jun. 1, 2009 is incorporated as a reference. 
     While the invention has been explained with reference to the specific embodiments of the invention, the explanation is illustrative and the invention is limited only by the appended claims.