Abstract:
A high data rate millimeter wave radio adapted to receive an binary input data at an input data rate in excess of 3.5 Gbps and to transmit at a transmit data rate in excess of 3.5 Gbps utilizing encoded three-bit data symbols on a millimeter carrier wave at a millimeter wave nominal carrier frequency, defining a carrier wavelength and period, in excess of 70 GHz with differential phase-shift keying utilizing eight separate phase shifts. Preferred embodiments of the invention can support many of the high data rate standards including the following group of protocols or standards: SONET OC-96 (4.976 Gbps); 4xGig-E (5.00 Gbps); 5xGig-E (6.25 Gbps); OBSAI RP3-01 (6.144 Gbps); 6xGig-E (7.50 Gbps); Fibre Channel 8GFC (8.5 Gbps); SONET OC-192 (9.952 Gbps); Fibre Channel 10GFC Serial (10.52 Gbps) and 10 GigaBit Ethernet.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application is a continuation in part of U.S. patent application Ser. No. 12/228,114, filed Aug. 7, 2008 which is incorporated herein by reference. 
    
    
     FIELD OF THE INVENTION 
     The present invention relates to radio systems and in particular to millimeter wave radio systems. 
     BACKGROUND OF THE INVENTION 
     Millimeter Wave Radios 
     In January of 2003, the United States Federal Communications Commission (with encouragement from Applicants&#39; employer and others) affected a new rulemaking in which 10 GHz of bandwidth, comprising (E-Band) frequency channels spanning the extremely high frequency bands from 71 to 76 gigahertz (GHz), 81 to 86 GHz and 92 to 95 GHz. At least 99 percent of the output power should be within these limits which are sometime referred to as the occupied transmit bandwidths. These bands are being used for short range, high bandwidth communications. These bands were opened for point-to-point radio communications without restrictions on modulation efficiency. The unique characteristic of this rulemaking was that for the first time a large section of bandwidth would be shared based on geographical (antenna beamwidth) constraints rather than spectral (channelizing) constraints. Each user of the new frequency bands was free to use the entire available bandwidth (5 GHz in each of two bands) as long as the transmit/receive path was confined within a single very narrow channel (“pencil beam” less than 1.2 degrees wide) in 3-dimensional space. However, rain fade at E Band can reach up to 35 dB/km, versus 1 dB/km at 6 GHz, so much larger link margins must be maintained at E Band, relative to lower frequency microwave bands, to accommodate severe weather events. On the other hand, the 5 GHz channel bandwidths available at E-Band are more than twenty times as wide as the widest channels available at microwave frequencies, so data rates in the range of 1 Gbps (unheard of for microwave radios) are made possible using simple on-off keying or binary phase-shift keying. Applicants&#39; employer led the way in the development of the early millimeter wave radios. Several patents describing these early radios have been awarded to Applicants and their fellow workers and assigned to Applicants&#39; employer. These patents include the following patents all of which are incorporated herein by reference: 
     
       
         
               
               
               
             
           
               
                   
               
               
                 PATENT 
                   
                   
               
               
                 NUMBER 
                 ISSUED 
                 TITLE 
               
               
                   
               
             
             
               
                 U.S. Pat. No. 6,556,836 
                 Apr. 29, 2003 
                 POINT TO POINT MILLLIMETER WAVE DUAL BAND FREE SPACE 
               
               
                   
                   
                 GIGABIT PER SECOND COMMUNICATION LINK 
               
               
                 U.S. Pat. No. 6,665,546 
                 Dec. 16, 2003 
                 HIGH SPEED POINT-TO-POINT MILLIMETER WAVE DATA 
               
               
                   
                   
                 COMMUNICATION SYSTEM 
               
               
                 U.S. Pat. No. 6,714,800 
                 Mar. 30, 2004 
                 CELLULAR TELEPHONE SYSTEM WITH FREE SPACE 
               
               
                   
                   
                 MILLIMETER WAVE TRUNK LINE 
               
               
                 U.S. Pat. No. 7,062,283 
                 Jun. 13, 2006 
                 CELLULAR TELEPHONE SYSTEM WITH FREE SPACE MILLIMETER 
               
               
                   
                   
                 WAVE TRUNK LINE 
               
               
                 U.S. Pat. No. 7,065,326 
                 Jun. 20, 2006 
                 MILLIMETER WAVE COMMUNICATIONS SYSTEM WITH A HIGH 
               
               
                   
                   
                 PERFORMANCE MODULAR CIRCUIT 
               
               
                 U.S. Pat. No. 7,680,516 
                 Mar. 16, 2010 
                 MOBILE MILLIMETER WAVE COMMUNICTION LINK 
               
               
                 U.S. Pat. No. 7,769,347 
                 08/03/2-10 
                 WIRELESS COMMUNICATION SYSTEM 
               
               
                   
               
             
          
         
       
     
     Quadrature Amplitude Modulation 
     Two key conditions which dictated early microwave radio designs for fixed point-to-point communications were: 1) very limited available bandwidth in which to transmit as much data as possible, and 2) minimal dynamic range variation associated with rain fade and other weather or atmospheric variations. The first of these conditions dictated the adoption of very high-order modulation techniques such as quadrature amplitude modulation (QAM). For example, 64-QAM, 128-QAM, or 256-QAM systems are in use for microwave links in which several (such as 6 to 8) digital bits can be sent simultaneously using a single pseudo-digital symbol, thereby increasing spectral throughput (bits per second per Hertz), but using such modulation schemes, a penalty of 17 to 22 dB in transmitter power is incurred relative to single-bit symbol modulation (on-off keying or binary phase shift keying) to maintain a manageable bit error rate. The second key condition driving earlier designs, the comparatively lower atmospheric attenuation and weather fade characteristic of lower frequencies, made such modulation efficiency trades desirable for microwave radios, by requiring little additional link margin to cope with high humidity and heavy rainfall. These QAM techniques work well on microwave systems, but have not been successfully applied to millimeter wave communication systems designed for longer propagation paths (e.g. exceeding a kilometer). 
     Modulation Using Phase Shift Keying 
     Phase shift keying is a digital modulation scheme that conveys information by changing, or modulating, the phase of a reference signal. Usually, each phase encodes an equal number of bits. Each pattern of bits forms a symbol that is represented by the particular phase. A demodulator which is designed specifically for the symbol-set used by the modulator, determines the phase of the received signal and maps it back to the symbol it represents, thus recovering the original data. This requires the receiver to be able to compare the phase of the received signal to a reference signal. Alternatively, instead of using the data bit pattern to set the absolute phase of the wave, it can instead be used to change the phase by a specified amount. The demodulator then determines the change in the phase of the received signal rather than the phase itself. Since this scheme depends on the difference between successive phases, it is termed differential phase-shift keying (DPSK). Differential phase shift keying can be significantly simpler to implement than ordinary phase shift keying since there is no need for the demodulator to have a copy of the reference signal to determine the exact phase of the received signal. 
     Costas Loops 
     A Costas loop is a phase locked loop used for carrier phase recovery from carrier modulation signals, such as from double-sideband suppressed carrier signals. It was invented by in the 1950s. The primary application of Costas loops is in wireless receivers. Its advantage over other similar phase detectors is that at small phase deviations the Costas loop error voltage is larger. This translates to higher sensitivity and also makes the Costas loop uniquely suited for tracking Doppler shifted carrier signals. 
     Last Mile and Middle Mile Communication Services 
     The United States and many other countries are crisscrossed by many thousands of miles of fiber optic communications links providing almost unlimited communication between major population centers. Telephone companies provide communications services to nearly all of the homes and offices in the United States and many other countries, but existing telephone services in many areas provide only low speed (i.e. low data rate) connections. Communication companies are rapidly improving these last mile services with cable and fiber optic connections, but these improvements are expensive and a large number of people are still without access to high speed communication services. Microwave radios have been used for many years for last mile and middle mile communication services, but bandwidths for these systems are typically limited such that data rates available are typically much less than 1 Gbps. Communication companies are beginning to utilize millimeter wave radios to provide these services but the data rates of these radios, although much greater than the microwave systems, are currently limited to about 1 Gbps. Many cellular systems are becoming overloaded due to the increased bandwidth required by the iPhone 4 and similar consumer products and prior art backhaul facilities are fast becoming inadequate. Updating these systems is expensive. 
     High Data Rate Protocols 
     A popular communication protocol which is being increasingly utilized to meet this demand for increased bandwidth is the Internet Protocol (IP) Ethernet Standard at 10 Gigabits per second, with a small amount of overhead for ensuring carrier modulation (bit toggling) at some minimum speed. There are, however, many current wired and fiber communications standards which involve serial transfer of binary data at speeds in excess of 4 Gigabits per second and within the maximum bandwidth capability of this radio. Some of these include: SONET OC-96 (4.976 Gbps); 4xGig-E (5.00 Gbps); 5xGig-E (6.25 Gbps); OBSAI RP3-01 (6.144 Gbps); 6xGig-E (7.50 Gbps); Fibre Channel 8GFC (8.5 Gbps); SONET OC-192 (9.952 Gbps) and Fibre Channel 10GFC Serial (10.52 Gbps) 
     What is needed is a high data rate millimeter wave radio providing data rates substantially greater than the data rates of existing millimeter wave radios, appropriately designed to deal with the added complexities associated with these higher frequencies contemplated by these high data rate protocols, and configured to take advantage of the much higher bandwidth available in the frequency ranges above 70 GHz. 
     SUMMARY OF THE INVENTION 
     The present invention provides a high data rate millimeter wave radio designed to receive binary input data at an input data rate in excess of 3.5 Gbps and to transmit at a transmit data rate in excess of 3.5 Gbps utilizing encoded three-bit data symbols on a millimeter carrier wave at a millimeter wave nominal carrier frequency in excess of 70 GHz. The radio uses differential phase-shift keying utilizing eight separate phase shifts. Embodiments of the invention can support many of the high data rate standards including the following group of protocols or standards: SONET OC-96 (4.976 Gbps); 4xGig-E (5.00 Gbps); 5xGig-E (6.25 Gbps); OBSAI RP3-01 (6.144 Gbps); 6xGig-E (7.50 Gbps); Fibre Channel 8GFC (8.5 Gbps); SONET OC-192 (9.952 Gbps); 10GigE (10.3125 Gbps) and Fibre Channel 10GFC Serial (10.52 Gbps). 
     A 10 Gigabit Per Second Millimeter Wave Radio 
     Preferred embodiments of the present invention include a 10-gigabits-per-second radio transceiver operating with carrier signals in the frequency ranges of 71-76 GHz and 81-86 GHz. This transceiver includes a transmitter transmitting in the 71-76 GHz band and a receiver receiving in the 81-86 GHz band, or a transmitter transmitting in the 81-86 GHz band and a receiver receiving in the 71-76 GHz band. Two of these transceivers constitute a millimeter-wave radio link designed to operate in accordance with the 10 Gigabit Ethernet standard (which is also referred to as 10GE, GbE and 10 GigE). In these preferred embodiments the transmitter for each transceiver supports a digital data rate of 10.3125 Gbps (10 Gigabits raw data per second plus IEEE 802.3 Clause 49 64b/66b encoding which accounts for the 0.3125 Gbps excess), using 3-bit symbol encoding (e.g. 8PSK). The carrier phase is modulated at a symbol rate of 3.4375 billion-symbols-per-second, so as to fit easily into the 5 GHz channel modulation bandwidth allowed by the prevailing FCC band plan for E-Band communication. During each (approximately 291 ps) symbol period, 3 bits of data are clocked into a temporary data buffer and then loaded onto three parallel data lines to form a most-significant-bit (MSB), a second-most-significant-bit (2SB) and a least-significant-bit (LSB) in a phase modulator. A nearly-instantaneous phase shift is imposed onto the transmitted carrier signal, each shift representing a specific digital symbol. This phase shift comprises one of eight standard phase shifts, with the degree of shift depending on the states of the three data lines. In a preferred embodiment the phase shift is either 0°, 45°, 9°, 135°, 180°, 225°, 270° or 315°. In this preferred embodiment each of these phase shifts respectively represent a symbol consisting of a combination of binary bits, 000 to 111, as shown in the following table: 
     
       
         
               
               
               
             
           
               
                   
                   
               
             
             
               
                   
                 0° 
                 000 
               
               
                   
                 45° 
                 001 
               
               
                   
                 90° 
                 010 
               
               
                   
                 135° 
                 011 
               
               
                   
                 180° 
                 100 
               
               
                   
                 225° 
                 101 
               
               
                   
                 270° 
                 110 
               
               
                   
                 315° 
                 111 
               
               
                   
                   
               
             
          
         
       
     
     In the preferred embodiments the receiver in each transceiver includes a demodulator with a sequential state phase comparator that detects and evaluates the received signal to reconstruct the three data bits from each symbol. 
     Preferred embodiments are adapted to receive an binary input data stream at an input data rate in excess of 3.5 Gbps and to transmit at a transmit data rate in excess of 3.5 Gbps utilizing encoded three-bit data symbols, on a carrier at a millimeter wave nominal carrier frequency in excess of 70 GHz, with differential phase-shift keying utilizing eight separate phase shifts at a phase-shift rate in excess of 1.16 billion phase shifts per second. To do this, the transmitter includes a frequency stabilized millimeter wave source operating at the millimeter wave carrier frequency; an encoder having an output clocked at said on-off keyed input data rate divided by three and adapted to generate sets of four phase shift modulator control bits, each set representing one of eight three-bit symbols; and a modulator adapted to apply each set of four phase shift modulator control bits from the output of the encoder to the millimeter carrier wave in the form of a single phase shift for each three-bit symbol, each phase shift being one of eight standard, recognizably distinct phase shifts. The receiver is adapted to receive an incoming millimeter wave signal transmitted from a remote millimeter wave transmitter transmitting at frequencies in excess of 70 GHz, said receiver intended to reconstruct communications data sent from the remote transmitter. The receiver includes a millimeter wave amplifier adapted to amplify said incoming millimeter wave signal and a demodulator adapted to decode the incoming millimeter wave signal to produce a binary output data stream at an output data rate in excess of 3.5 Gbps. 
     In preferred embodiments operating at data rates in the range of about 3.5 Gbps the occupied transmit bandwidth is preferably between 1.0 GHz and 5 GHz. For data rates of the 10 GigE transceiver the occupied transmit bandwidth is preferably between 3.5 GHz and 5 GHz. Preferably the power spectrum density within more than 70 percent of the output power of the transmitter is constant to within +/−1.5 dB and the transceiver provides provide a dynamic range in power output exceeding 15 dB. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows features to two radios for a preferred embodiment of a 10.3125 Gbps millimeter radio link. 
         FIGS. 1A and 1B  are drawings showing features of a modulator for a first preferred embodiment of the present invention. 
         FIGS. 2A and 2B  are drawings showing features of a modulator for a second preferred embodiment of the present invention. 
         FIG. 3  shows important components of a D8PSK state adder. 
         FIG. 4  describes drive logic for a preferred embodiment of the present invention. 
         FIG. 5  is a state table applicable to a preferred embodiment. 
         FIG. 6  is a demodulator circuit for preferred embodiments. 
         FIG. 6A  shows features of an 8PSK Costas loop for stabilizing the  FIG. 6  circuit. 
         FIG. 7  is a drawing from the parent application showing features of the radios described therein. 
     
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     First Preferred Embodiment 
     Applicants&#39; 10-GigE Radio 
     A first preferred embodiment of the present invention satisfying the needs outlined above is a 10-GigE radio. The radio is based on and is a substantial improvement of the basic circuit design of a 3.072 Gbps radio described in the parent application, of which the present application is a continuation in part. That application is U.S. patent application Ser. No. 12/228,114, filed Aug. 7, 2008, Millimeter Wave Radio with Phase Modulation, which is incorporated herein by reference. This parent radio utilized radio circuitry as specifically described in  FIG. 3  of the above patent which is reproduced in this application as  FIG. 7 . 
     To support a digital data rate of 10.3125 Gbps (10 Gigabits raw data per second plus IEEE 802.3 Clause 49 64b/66b encoding which accounts for the excess 0.3125 Gbps), the radio uses 3-bit symbol encoding and has its carrier modulated at a symbol rate of 3.4375 Giga-symbols-per-second so as to fit into the 5 GHz channel modulation bandwidth allowed by the prevailing FCC band plan for E-Band communication. The product of the 3.4375 billion symbols per second and the 3 bits per symbol results in the above digital data rate of 10.3125 billion bits per second. 
     In preferred embodiments the millimeter wave radios are marketed as a pair of radios making a 10-GigE millimeter radio link. Details showing the important electronic components are described by reference to  FIGS. 1A and 1B . One of the radios, Radio A, transmits at an average frequency of 73.5 GHz with a permitted total bandwidth of 5 GHz and receives at an average frequency of 83.5 GHz, and the other radio, radio B, transmits at an average frequency of 83.5 GHz with a permitted total bandwidth of 5 GHz and receives at an average frequency of 73.5 GHz. Components of radio A are identified with reference numbers between  100  and  140  and components of radio B are identified with reference numbers between  200  and  240 . 
     Transmitter Circuits 
     As shown in  FIG. 1  a microwave signal is generated, using phase locked voltage controlled oscillators  110  and  210 , chosen for very low phase noise. These oscillators, such as Hittite Part Number HMC783 and Part Number HMC807, are locked (at 12.25 GHz for Radio A and 13.92 GHz for Radio B respectively) to a crystal reference. These microwave signals are amplified in amplifiers  111  and  211 , frequency doubled to 24.5 GHz and 27.83 GHz (in multipliers  112  and  212 ) filtered in filters  113  and  213 , amplified in amplifiers  115  and  215 , and then frequency tripled in times-three frequency multipliers  114  and  214  (to frequencies of 73.5 GHz and 83.5 GHz respectively), to produce carrier frequencies preferably approximately centered in the 71-76 band (73.5 GHz for Radio A) and 81-86 GHz band (83.5 GHz for Radio B), with resulting integrated RF phase noise of less 5 degrees. This signal passes to the phase-shift keyed modulators  118  and  218  and on to a power amplifier  120  and  220 , for output to the antenna diplexer  122  for Radio A and diplexer  222  for Radio B. Key improvements over the radio of the parent application and the earlier prior art is encompassed in modulators  118  and  218  and are described below with reference to  FIG. 1A . 
     The transmitter power amplifier  120  and  220  includes at least 15 dB of power adjustment by way of a control voltage to the transmitter power amplifiers  120  and  220 , designed to compensate for rain fade by way of an in-band or out-of-band handshake communication of power margin with a remote receiver. The full amplifier chain is designed to provide less than 3 dB of gain variation across the full (approximately 4 GHz) operating bandwidth of the transmitter. The power amplifiers  120  and  220  are designed for high efficiency and their saturation power will not typically exceed 200 milliwatts. Any requirement for higher output power can be accommodated by a separate external amplifier between the transceiver chip and the diplexer. 
     Modulation of the Transmit Carrier Signal 
     D8PSK Modulator 
     Important components of modulators  118  and  218  are shown at  18  in  FIG. 1A  for a preferred modulator design. This modulator supports a digital data rate of 10.3125 Gbps (10 Gigabits raw data per second plus IEEE 802.3 Clause 49 64b/66b encoding), using 3-bit symbol encoding (e.g. 8PSK) to modulate the carrier signals at a symbol rate of 3.4375 Giga-symbols-per-second (so as to fit into the 5 GHz channel modulation bandwidth allowed by the prevailing FCC band plan for E-Band radios). Modulator  18  includes input signal splitter  352 , output signal combiner  354 , 45-degree phase delay line  350 , and two IQ modulators  318 A and  318 B. The IQ modulators, such as Göteborg Microwave Integrated Circuits model MDR080A01, incorporate 90-degree hybrid splitters  346  and  348 , phase conserving/inverting mixers  344 A,  344 B,  344 C and  344 D, and combiners  356  and  358 . Splitter  352  divides the input signal into two equal signals a reference signal in line  353  and a second signal in line  354 . The signal in line  354  is phase shifted by 45 degrees by delay line  350 . In IQ modulator  318 B this signal is then split through 90-degree hybrid  348  into two parts which are phase shifted respectively by a total of 45 degrees and 135 degrees relative to the initial reference signal, and these signals are passed through two driver-controlled switchable phase inverters before being recombined at  345 B. In IQ modulator  318 A, the reference signal is split into two parts by 90-degree hybrid  346 , one part of which is not phase shifted and the other part of which is phase shifted 90 degrees, and then passed through two driver-controlled switchable phase inverters before being recombined at  345 A. The signals from IQ modulators  318 A and  318 B are then further combined in combiner  354  to form an 8PSK-modulated waveform. 
     In this preferred embodiment modulator  18  is controlled to shift the phase of the carrier signal (either 73.5 GHz or 83.5 GHz) at approximately 291 ps intervals (a rate of 3.4375 billion shifts per second). The amount of shift is one of the following: 0°, 45°, 90°, 135°, 180°, 225°, 270° or 315°. The carrier signal enters the modulator as shown at  340  with an incoming phase and exits the modulator as shown at  342  with an exit phase. The shifts are imposed on the 73.5 GHz or 83.5 GHz carrier signal with the 8PSK modulator shown in  FIG. 1A  using drive signals A, B, C and D as shown in  FIG. 1A . The logic state of each of drivers is either +1 or −1; the preferred IQ modulators use differential line pairs to deliver these drive states to the mixers. Simple drivers at A, B, C, and D output signals with positive polarity, which causes the mixers  344 A,  344 B,  344 C and  344 D to preserve signal polarity, or signals with negative polarity, causing the mixers to invert signal polarity. 
       FIGS. 3 ,  4  and  5  explains how phase control bits are encoded in encoders  116  and  216  (shown in  FIG. 1 ) onto the 73.5 GHz and the 83.5 GHz carrier signals once each 291 ps, in the form of a single phase shift selected from one of the eight recognizably distinct phases listed in the State Table in  FIG. 5 . In the encoders, data are clocked at the nominal 10-Gigabit-Ethernet rate of 10.3125 Gbps into a 3-bit temporary buffer, and then transferred simultaneously as a three-bit symbol at 3.4375 (equal to 10.3125 divided by 3) billion symbols per second onto three parallel data lines, identified as a most-significant bit (Data MSB), second-most-significant bit (Data 2SB), and least-significant bit (Data LSB). These data are input to a DPSK State Adder ( FIG. 3 ), where they are used to compute a three-bit control word to feed the Drive Logic ( FIG. 4 ) which in turn drives the D8PSK modulator ( FIG. 1B ). 
     Encoders 
     Phase control bits are generated using a summing circuit in each of encoders  116  and  216  shown in  FIG. 1 . Each of these summing circuits is referred to as a D8PSK State Adder and is shown in  FIG. 3 . This summing circuit is used to generate eight phase shift keyed modulator control bits from the 10-GigE data stream  103  and  203  received by the receive encoders  116  and  216  as shown in  FIG. 1 . In this summing circuit the current data least significant bit (Data LSB i-1 ) is added to the control least significant bit (Control LSB) from the prior (291 ps earlier) symbol of three bits, as shown at  322 . The same is done for the second significant bit and the most significant bit as shown at  324  and  326  utilizing XOR, AND, and OR logic circuits, also as shown in  FIG. 3 . The carry bit from the sum of the most significant bits is truncated. The result is the creation of current Control LSB i , Control 2SB i ; and Control MSB i ; as shown at  328 . Thus, the input to the summing circuit is the three-bit phase change instruction data as shown in  FIG. 3  and the output is the three-bit absolute phase control instruction. 
     Drive Logic 
     The State Table in  FIG. 5  provides eight phase control bit patterns, 000, 001, 010, 011, 100, 101, 110 and 111, and eight sets of corresponding modulator inputs for each of the four modulator drive signals, A, B, C and D, for the 8PSK transmitter modulator  18  shown in detail in  FIG. 1A . As described previously, the phase control bits shown in the State Table do not represent the data stream itself, but rather the sum of the three current data bits added to the prior phase control bits and truncated to the three least significant bits and discarding any fourth bit. As such the transmitted data is encoded into the phase shift rather than the phase, so that no absolute phase calibration is necessary at the remote receiver. The IQ modulator inputs A, B, C, and D are encoded with the 8PSK modulator drive logic using the XOR, AND, and inverter logic circuits as shown in  FIG. 4 . The 8PSK modulator drive logic, generates drive signals for the IQ modulators  318 A and  318 B shown in  FIG. 1A . The drive signals are according to the phase encoding plan shown in the State Table in  FIG. 5 . Simple drivers at the output generate positive polarity, causing the mixers to preserve signal polarity, for a logic state of +1 as shown in the State Table in  FIG. 5 , or generate negative polarity, causing the mixers to invert signal polarity for a logic state of zero (represented by −1 in the State Table  FIG. 5 ). 
     8PSK Modulator Phase Diagram 
     As explained above and as shown in  FIG. 1A  signals entering phase conserving/inverting mixers  344 A,  344 B,  344 C and  344 D are offset relative to each other; i.e. The signal entering mixer  344 A is 0 degrees,  344 B is 90 degrees,  344 C is 45 degrees and  344 D is 135 degrees. So, for example, to transmit a three-bit symbol representing the three-bit symbol 010, modulator  18  would impose a phase shift of 157.5 degrees as shown at  360  in  FIG. 5 . As shown in the State Table of  FIG. 5  this calls for an A drive signal of −1, a B drive signal of +1, a C drive signal of −1 and a D drive signal of +1. The −1 A-Drive signal causes inverter  344 A to switch its 0 degree signal to 180 degrees; the +1 B-Drive signal causes inverter  344 B to retain its 90 degree signal; the −1 C-Drive signal causes inverter  344 C to switch its 45 degree (45 degrees plus 0 degrees) signal to 225 degrees and the +1 D Drive signal causes inverter  344 D to retain its 135 degree (45 degrees plus 90 degrees) signal at 135 degrees. The net effect is a net shift of 157.5 degrees as shown by the dashed vector line  362  on the 8PSK Modulation Phase Diagram shown in  FIG. 1B . 
     Receiver Circuits 
     The principal receiver circuits for Radio A and Radio B are shown in  FIG. 1 . They include low noise amplifier  124  and  224  and demodulator  126  and  226 . Incoming millimeter wave signals at nominal frequencies of 73.5 GHz or 83.5 GHz are collected in antennas  140  and  240 , separated in diplexer  122  and  222 , amplified in low noise amplifier  124  and  224 , filtered in band pass filter  125  and  225  to filter out out-of-band noise, and demodulated in demodulator  126  and  226  where they may drive a fiber-optic transmitter or other transmission medium to produce a 10.3125 Gbps on-off data output stream as shown at  105  and  205  in  FIG. 1 . 
     D8PSK Demodulator 
     Details of the demodulation circuits are described in  FIG. 6  and  FIG. 7 . In these circuits the differential 8-state phase shift is detected and evaluated using a sequential state phase comparator, to reconstruct the most, second and least significant bits, MSB, 2SB, and LSB of each of the three-bit symbols at the rate of 10.3125 Gbps. As shown in  FIG. 6  delay line  400  corresponding to an integral number of carrier wavelengths, matching as closely as possible the total phase change across one symbol period (291 ps), and a 45 degree delay line and a pair of 90 degree hybrids are used to compare the phase state of the current data symbol with the previous data symbol. 
     The current signal is input as shown at  390  and split in splitter  392  into a current state signal  394  and a previous state signal  396 . One-half of the input signal  390  is directed through the delay line  400  (delay time matching integral wave periods closest to one 291 ps symbol period; i.e. 285.7 ps [exactly 21 periods] for a 73.5 GHz demodulator, and 287.4 ps [exactly 24 periods] for a 83.5 GHz demodulator) to create the previous signal  396 . A second λ/8 (corresponding to 45 degrees) delay line  402  and a pair of IQ modulators  418 A and  418 B (such as Göteborg Microwave Integrated Circuits model MDR080A01) consisting of 90-degree hybrids  404  and  406 , are used to offset the phase of the current signal from the preceding signal by zero, π/4, π/2 and 3π/4 radians (0 degrees, 45 degrees 90 degrees and 135 degrees). Signal splitters  424  and  426 , and mixers  410 ,  412 ,  420  and  422 ) are used to mix the current and delayed signals with the aforementioned phase shifts into four output channels, A, B, C, and D. The preferred IQ modulator outputs each of these signals through a filter  414  on to differential line pairs (pair A and C and pair B and D) with very low noise. The demodulator is designed to be insensitive to the amplitudes of the intermediate mixing states, detecting the location and polarities of the unbalanced (±1, 0) and balanced (±√2/2, ±√2/2) output pairs of the IQ modulators to decode the three-bit data from the differential phase shift. The demodulator State Table corresponding to the  FIG. 6  demodulator is shown in Table 1. In Table 1 entries of “s” represent nominal relative amplitudes of √2/2. Amplitudes of voltage signals at nodes marked A, B, C, D, X, Y, LSB, 2SB and MSB in  FIG. 6  are tracked for each differential phase shift state and are utilized to determine the values of the three bits of each symbol, at the rate of 3.4375 billion symbols per second. 
     The long delay line  400  shown in  FIG. 6  used in the sequential state comparator is approximately one symbol period (291 ps) long and chosen to match an exact number of wave periods of the carrier; for instance exactly 21 cycles of a carrier at 73.5 GHz and exactly 24 cycles of a carrier at 83.5 GHz. This corresponds to between 2 and 3 inches of stripline or microstrip transmission line. Stabilization of this path length can be accomplished, using an error signal from a Costas-type loop as described below to provide a small incremental feed-back correction path or alternatively by varying the mixer bias levels slightly to affect the required phase adjustment. The 45 degree increment of ambiguity in the 8PSK Costas-type loop is large compared to the delay length variation that can be expected from electrical or thermo-mechanical effects, so the delay line should be adequately stabilized in all conditions. 
     
       
         
               
               
               
               
               
               
               
               
               
               
               
             
               
               
               
               
               
               
               
               
               
               
               
             
           
               
                 TABLE 1 
               
               
                   
               
               
                 Phase 
                   
                   
                   
                   
                   
                 X = 
                 Y = 
                 sgn(A + B + 
                 −sgn(X + 
                 sgn(Y 2  − 
               
               
                 Shift 
                 Data 
                 A 
                 B 
                 C 
                 D 
                 A 2  − C 2   
                 B 2  − D 2   
                 C + D) = MSB 
                 Y) = 2SB 
                 X 2 ) = LSB 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                  0° 
                 000 
                 1 
                   s 
                 0 
                 −s 
                 1 
                 0 
                 −1 (0) 
                 −1 (0) 
                 −1 (0) 
               
               
                  45° 
                 001 
                   s 
                 1 
                   s 
                 0 
                 0 
                 1 
                 −1 (0) 
                 −1 (0) 
                 1 
               
               
                  90° 
                 010 
                 0 
                   s 
                 1 
                   s 
                 −1 
                 0 
                 −1 (0) 
                 1 
                 −1 (0) 
               
               
                 135° 
                 011 
                 −s 
                 0 
                   s 
                 1 
                 0 
                 −1 
                 −1 (0) 
                 1 
                 1 
               
               
                 180° 
                 100 
                 −1 
                 −s 
                 0 
                   s 
                 1 
                 0 
                 1 
                 −1 (0) 
                 −1 (0) 
               
               
                 225° 
                 101 
                 −s 
                 −1 
                 −s 
                 0 
                 0 
                 1 
                 1 
                 −1 (0) 
                 1 
               
               
                 270° 
                 110 
                 0 
                 −s 
                 −1 
                 −s 
                 −1 
                 0 
                 1 
                 1 
                 −1 (0) 
               
               
                 315° 
                 111 
                   s 
                 0 
                 −s 
                 −1 
                 0 
                 −1 
                 1 
                 1 
                 1 
               
               
                   
               
             
          
         
       
     
     8PSK Costas Loop 
     These 8PSK Costas-type loops exists in prior art. The one designed for this preferred embodiment is shown in  FIG. 6A . It is used in this preferred embodiment to stabilize the path delay as an exact integral number of wavelengths. The modulated communication waveform can be mathematically described as: 
               S   =       a   ⁢           ⁢     cos   ⁡     (     ω   ⁢           ⁢   t     )         +     b   ⁢           ⁢     cos   ⁡     (       ω   ⁢           ⁢   t     +     π   4       )         +     c   ⁢           ⁢     cos   ⁡     (       ω   ⁢           ⁢   t     +     π   2       )         +     d   ⁢           ⁢     cos   ⁡     (       ω   ⁢           ⁢   t     +       3   ⁢   π     4       )             ,         
where the eight defined 8PSK symbol states are represented by:
 
     
       
         
               
               
               
               
               
             
               
               
               
               
               
             
           
               
                   
               
               
                 Symbol 
                 a 
                 b 
                 c 
                 d 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 000 
                 1 
                 0 
                 0 
                 0 
               
               
                 001 
                 0 
                 1 
                 0 
                 0 
               
               
                 010 
                 0 
                 0 
                 1 
                 0 
               
               
                 011 
                 0 
                 0 
                 0 
                 1 
               
               
                 100 
                 −1 
                 0 
                 0 
                 0 
               
               
                 101 
                 0 
                 −1 
                 0 
                 0 
               
               
                 110 
                 0 
                 0 
                 −1 
                 0 
               
               
                 111 
                 0 
                 0 
                 0 
                 −1 
               
               
                   
               
             
          
         
       
     
     In the receiver IQ modulator shown in  FIG. 6 , the received signal  406  is mixed with the reference signal  408  which has been delayed by an exact multiple of the carrier wave period plus some small error amount δ; the Costas loop is designed to minimize this error. After a short transient interval at the beginning of each symbol period (the difference between the symbol period of 290.9 ps and the delay line period of 285.7 or 287.4 ps), the two inputs to the mixer represent two successive bit phases offset by the encoded phase shift plus the small error amount. In order to minimize this error, the Costas loop uses as one input the modulated signal waveform corresponding to the present symbol state, and as the second input the delayed (previous) symbol state, defining the phase shift of the reference signal to be zero plus the small error:
 
 R =cos(ω t +δ),
 
     Multiplying the received signal by the reference signal into the A and C channels, with and without a π/2 phase shift respectively, and filtering the frequency doubled components, 
                         A   =       ⁢         a   2     ⁢   cos   ⁢           ⁢   δ     +       b   2     ⁢     cos   ⁡     (       π   4     +   δ     )         +                     ⁢         c   2     ⁢     cos   ⁡     (       π   2     +   δ     )         +       d   2     ⁢     cos   ⁡     (         3   ⁢   π     4     +   δ     )                       =       ⁢       1   2     ⁡     [             cos   ⁢           ⁢   δ   ⁢     {     a   +         2     2     ⁢   b     -         2     2     ⁢   d       }       -               sin   ⁢           ⁢   δ   ⁢     {           2     2     ⁢   b     +   c   +         2     2     ⁢   d       }             ]                     =       ⁢       1   2     ⁡     [       I   ⁢           ⁢   cos   ⁢           ⁢   δ     -     Q   ⁢           ⁢   sin   ⁢           ⁢   δ       ]         ,                 [       Node   ⁢           ⁢   1     ,     shown   ⁢           ⁢   in   ⁢           ⁢   box   ⁢           ⁢   in   ⁢           ⁢     FIG   .           ⁢   6     ⁢   A       ]                     C   =       ⁢         a   2     ⁢     cos   ⁡     (       π   2     +   δ     )         +       b   2     ⁢     cos   ⁡     (         3   ⁢   π     4     +   δ     )         +                     ⁢         c   2     ⁢     cos   ⁡     (     π   +   δ     )         +       d   2     ⁢     cos   ⁡     (         5   ⁢   π     4     +   δ     )                       =       ⁢     -       1   2     ⁡     [             cos   ⁢           ⁢   δ   ⁢     {           2     2     ⁢   b     +   c   +         2     2     ⁢   d       }       +               sin   ⁢           ⁢   δ   ⁢     {     a   +         2     2     ⁢   b     -         2     2     ⁢   d       }             ]                       =       ⁢     -       1   2     ⁡     [       Q   ⁢           ⁢   cos   ⁢           ⁢   δ     +     I   ⁢           ⁢   sin   ⁢           ⁢   δ       ]           ,                 [       Node   ⁢           ⁢   2     ,     in   ⁢           ⁢     FIG   .           ⁢   6     ⁢   A       ]               
where we have defined
 
     
       
         
           
             
               I 
               ≡ 
               
                 a 
                 + 
                 
                   
                     
                       2 
                     
                     2 
                   
                   ⁢ 
                   b 
                 
                 - 
                 
                   
                     
                       2 
                     
                     2 
                   
                   ⁢ 
                   d 
                 
               
             
             ; 
             
               Q 
               ≡ 
               
                 
                   
                     
                       2 
                     
                     2 
                   
                   ⁢ 
                   b 
                 
                 + 
                 c 
                 + 
                 
                   
                     
                       2 
                     
                     2 
                   
                   ⁢ 
                   
                     d 
                     . 
                   
                 
               
             
           
         
       
     
     Analyzing the 8PSK Costas loop, the mixers, amplifiers and summers in the loop create the products 2AC(C 2 −A 2 ) and (C 2 −A 2 ) 2 −(2AC) 2  and then mix (multiply) these to create the phase error signal. Evaluating these products: 
                       2   ⁢   AC     =       1   4     ⁡     [         (       Q   2     -     I   2       )     ⁢   sin   ⁢           ⁢   2   ⁢   δ     -     2   ⁢   IQ   ⁢           ⁢   cos   ⁢           ⁢   2   ⁢   δ       ]         ,           [     Node   ⁢           ⁢   3     ]                     C   2     -     A   2       =       1   4     ⁡     [         (       Q   2     -     I   2       )     ⁢   cos   ⁢           ⁢   2   ⁢   δ     +     2   ⁢   IQ   ⁢           ⁢   sin   ⁢           ⁢   2   ⁢   δ       ]         ,           [     Node   ⁢           ⁢   4     ]                       2   ⁢     AC   ⁡     (       C   2     -     A   2       )         =       1   32     ⁡     [               {         (       Q   2     -     I   2       )     2     -       (     2   ⁢   IQ     )     2       }     ⁢   sin   ⁢           ⁢   4   ⁢   δ     -               4   ⁢     IQ   ⁡     (       Q   2     -     I   2       )       ⁢   cos   ⁢           ⁢   4   ⁢   δ           ]                     =       1   32     ⁡     [       Z   ⁢           ⁢   sin   ⁢           ⁢   4   ⁢   δ     -     W   ⁢           ⁢   cos   ⁢           ⁢   4   ⁢   δ       ]         ,                 [     Node   ⁢           ⁢   5     ]                           (       C   2     -     A   2       )     2     -       (     2   ⁢   AC     )     2       =       1   16     ⁡     [               {               (       Q   2     -     I   2       )     2     -                 (     2   ⁢   IQ     )     2           }     ⁢   cos   ⁢           ⁢   4   ⁢   δ     +               4   ⁢     IQ   ⁡     (       Q   2     -     I   2       )       ⁢   sin   ⁢           ⁢   4   ⁢   δ           ]                     =       1   16     ⁡     [       Z   ⁢           ⁢   cos   ⁢           ⁢   4   ⁢   δ     +     W   ⁢           ⁢   sin   ⁢           ⁢   4   ⁢   δ       ]         ,                 [     Node   ⁢           ⁢   6     ]               
where we have defined
 
 Z ≡( Q   2   −I   2 ) 2 −(2 IQ ) 2   ; W≡ 4 IQ ( Q   2   −I   2 ).
 
     The error signal is derived from the product of these last two terms: 
     
       
         
           
             
               
                 
                   
                     2 
                     ⁢ 
                     
                       
                         AC 
                         ⁡ 
                         
                           ( 
                           
                             
                               C 
                               2 
                             
                             - 
                             
                               A 
                               2 
                             
                           
                           ) 
                         
                       
                       ⁡ 
                       
                         [ 
                         
                           
                             
                               ( 
                               
                                 
                                   C 
                                   2 
                                 
                                 - 
                                 
                                   A 
                                   2 
                                 
                               
                               ) 
                             
                             2 
                           
                           - 
                           
                             
                               ( 
                               
                                 2 
                                 ⁢ 
                                 AC 
                               
                               ) 
                             
                             2 
                           
                         
                         ] 
                       
                     
                   
                   = 
                   
                     
                       
                         1 
                         1024 
                       
                       ⁡ 
                       
                         [ 
                         
                           
                             
                               ( 
                               
                                 
                                   Z 
                                   2 
                                 
                                 - 
                                 
                                   W 
                                   2 
                                 
                               
                               ) 
                             
                             ⁢ 
                             sin 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             8 
                             ⁢ 
                             δ 
                           
                           - 
                           
                             2 
                             ⁢ 
                             ZW 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             cos 
                             ⁢ 
                             
                                 
                             
                             ⁢ 
                             8 
                             ⁢ 
                             δ 
                           
                         
                         ] 
                       
                     
                     . 
                   
                 
               
               
                 
                   [ 
                   
                     Node 
                     ⁢ 
                     
                         
                     
                     ⁢ 
                     7 
                   
                   ] 
                 
               
             
           
         
       
     
     Evaluating this error signal for all eight 8PSK symbol states, the error signal is independent of the symbol state (and therefore is not modulated at the symbol rate): 
     
       
         
               
               
               
               
               
               
               
               
               
               
             
               
               
               
               
               
               
               
               
               
               
             
           
               
                   
               
               
                   
                   
                   
                   
                   
                   
                   
                   
                   
                 Node 7, 
               
               
                 Symbol 
                 a 
                 b 
                 c 
                 d 
                 I 
                 Q 
                 Z 
                 W 
                 Error Signal 
               
               
                   
               
             
             
               
                   
               
             
          
           
               
                 000 
                 1 
                 0 
                 0 
                 0 
                 1 
                 0 
                 1 
                 0 
                 sin(8δ)/1024 
               
               
                 001 
                 0 
                 1 
                 0 
                 0 
                  {square root over (2/2)} 
                  {square root over (2/2)} 
                 −1 
                 0 
                 sin(8δ)/1024 
               
               
                 010 
                 0 
                 0 
                 1 
                 0 
                 0 
                 1 
                 1 
                 0 
                 sin(8δ)/1024 
               
               
                 011 
                 0 
                 0 
                 0 
                 1 
                 −{square root over (2/2)} 
                  {square root over (2/2)} 
                 −1 
                 0 
                 sin(8δ)/1024 
               
               
                 100 
                 −1 
                 0 
                 0 
                 0 
                 −1 
                 0 
                 1 
                 0 
                 sin(8δ)/1024 
               
               
                 101 
                 0 
                 −1 
                 0 
                 0 
                 −{square root over (2/2)} 
                 −{square root over (2/2)} 
                 −1 
                 0 
                 sin(8δ)/1024 
               
               
                 110 
                 0 
                 0 
                 −1 
                 0 
                 0 
                 −1 
                 1 
                 0 
                 sin(8δ)/1024 
               
               
                 111 
                 0 
                 0 
                 0 
                 −1 
                  {square root over (2/2)} 
                 −{square root over (2/2)} 
                 −1 
                 0 
                 sin(8δ)/1024 
               
               
                   
               
             
          
         
       
     
     Design Considerations 
     Need for Flat Gain and Constant Group Delay 
     For these radios broad modulation/demodulation bandwidths will be carried through heterodyne or super-heterodyne up/down-conversion to result in transmitter/receiver front-end bandwidths covering up to 5 GHz of millimeter-wave frequency. As a result, millimeter-wave radio frequency transmitter and receiver components should exhibit flat gain (3 dB gain window) and constant group delay (&lt;50 ps) characteristics over a much broader bandwidth than the counterparts in the lower-frequency microwave radios. E-Band radio designs should feature amplifier and filter components with extremely low variation in group delay in order to faithfully preserve the superposition of spectral components that makes up a clean, bit-error free signal waveform (eye-diagram). 
     Need for Large Dynamic Range 
     Due to the severity of rain fade at E Band, the radio should operate over an expected dynamic range far in excess of a typical microwave radio to accommodate heavy rain events. The transmitter must have the capability for changing output power over a large dynamic range on command. 
     Transmitter Output Power 
     Ultimate output power is less important in E-Band radio than at lower frequencies, because rain fade quickly nullifies the benefits of a few dB of extra power even over a relatively short (approximately 1 km) link. Antenna gain is much higher at millimeter-wave—relative to microwave—for a given antenna size, so effective radiated power (ERP) is greatly enhanced by antennas of modest size, further reducing the importance of an expensive and reliability-limiting power amplifier in the transmitter. An optimal E-Band radio design will have a typical output power not exceeding 200 milliwatts, but with flat gain and phase characteristics across the full operating band of the radio (1 to 5 GHz) and allowing for a large dynamic range in output power. At frequencies above 70 GHz high humidity and heavy rainfall results in substantial increases in atmospheric attenuation, so any excess link margin at these frequencies is needed to cope with weather-related signal fade, rather than for increased modulation efficiency. 
     Need for Low Oscillator Phase Noise 
     The local oscillator used as a frequency source in the transmitter multiplier chain and/or heterodyne transmitter/receiver up-converters and down-converters should have extremely low phase noise (integrated double-sideband phase noise less than 1 degree at the microwave frequency of the oscillator; less than 5 degrees after frequency multiplications to E Band), in order to maintain an adequate spacing between phase states and thereby avoid oscillator-induced bit errors. 
     Need for Low Order Modulation 
     The strong atmospheric attenuation associated with rain events is accompanied by large temporal variations in the signal amplitude and phase received from a remote transmitter. This effect makes it difficult to distinguish small differences in amplitude and phase imposed by a modulator from those imposed by the atmosphere, leading to high bit error rates from radios using high-order modulation schemes. The most robust modulation schemes are on-off keying and binary phase shift keying (OOK and BPSK), which require at least 1 Hz of bandwidth for each bit-per-second of data throughput. This modulation efficiency is acceptable for E-Band radios supporting up to at least 3.072 Gbps of data throughput (OBSAI protocols). For radios supporting 10-Gigabit Ethernet (10Gig-E), the modulation efficiency must exceed 2 bits per second per Hz (e.g. 8PSK at 3 bits per second per Hz), but any higher order modulation schemes, typical of microwave radios, will be detrimental to radio performance. 
     Need for Amplitude-Insensitive Demodulator 
     The need for large power margin to accommodate rain events will often require the E-Band transmitter to transmit into the compression region of the output power amplifier. The symbol demodulator must be designed to be insensitive to amplitude, relying only on a power threshold and the polarity of the demodulated signals, so that the transmitter power amplifier may be pushed into compression during heavy rain events without significant degradation of symbol discrimination (bit error rate). 
     Output Power and Spectrum Gain Control 
     For this preferred embodiment shown in  FIG. 1  operating at data rates of the 10.3125 Gbps the occupied transmit bandwidth is between 3.5 GHz and 5 GHz. The transceiver is designed to provide a power spectral density, over more than 70 percent of occupied bandwidth of the transmitter which is constant to within +/−1.5 dB and the transceiver provides a dynamic range in power output exceeding 15 dB. 
     Variations 
     Although preferred embodiments of the present invention have been described in detail above, persons skilled in the radio art will recognize that many variations are possible within the scope of the present invention. Some variations are listed below. 
     Silicon-Germanium Semiconductors 
     The components of the millimeter wave radios described above are in general state of the art millimeter wave and optical fiber components. However, many of the components could be fabricated together on one or more semiconductor substrates to produce very low cost millimeter wave radios. Silicon-germanium bipolar transistors on complementary metal-oxide-semiconductor (referred to as “SiGe BiCMOS”) technology, which marries the superior low-noise and high-speed properties of the SiGe heterojunction bipolar transistors with the low cost and manufacturability advantages of conventional CMOS technology, represent an ideal solution for mixed-signal applications such as millimeter-wave wireless communications systems, in which frequency sources and multipliers, mixers and low-noise amplifiers are used alongside digital modulator control and processing circuitry. Amplifiers using SiGe bipolar transistors are more efficient and achieve lower noise figures than comparable conventional CMOS amplifiers, and the higher breakdown voltage of SiGe allows for higher device output power as well. 
     High Frequency Radio Components on Si and SiGe 
     Gallium Arsenide (GaAs) is superior to SiGe semiconductors for ultra-low phase noise high-frequency oscillators (so an external microwave phase-locked voltage-controlled oscillator (PLVCO) is a preferred frequency source), but the frequency multiplier chain, upconversion and downconversion mixers and millimeter-wave, microwave and baseband amplifiers can all be implemented satisfactorily using conventional microstrip circuitry on Si and SiGe semiconductor substrates. For lowest cost, a silicon wafer can be used as a substrate, with germanium placed locally on the chip at the locations of the millimeter-wave transistors and diodes, so that the more expensive SiGe material is localized only in the regions of the high-frequency MMW and microwave semiconductor junctions. Lower frequency circuitry, including the data encoder, high-speed driving logic and all other baseband and digital data and control electronics may be implemented using standard CMOS processes on the same silicon substrate. The data decoder and delay-stabilizing Costas loop can be implemented on the receiver chip, but due to its physical size the longer symbol-period delay line is best implemented off-chip, using a microstrip line on a flex-circuit material such as Rogers 3003 or using a waveguide delay line. A SiGe foundry, such as the IBM foundry located near Essex Junction, Vt., with a 0.13-micron or 0.09-micron SiGe process can produce SiGe chips of the preferred design for 10 Gbps E-Band radio transceivers. 
     Other Variations 
       FIGS. 2A and 2C  show features of a variation where the 45 degree delay line  350  shown in  FIG. 1A  is omitted. 
     The radio described in this patent is capable of delivering data rates in excess of 3.5 Gigabits per second, a speed which is unprecedented for any form of wireless communications at the time of this invention. The preferred embodiment in this description operates under the Internet Protocol (IP) Ethernet Standard at 10 Gigabits per second with a small amount of overhead for ensuring bit toggling at some minimum speed. There are, however, many other communications standards which involve serial transfer of binary data at speeds in excess of 3.5 Gigabits per second and within the maximum bandwidth capability of this radio. Some of these include: 
     SONET OC-96 (4.976 Gbps) 
     4xGig-E (5.00 Gbps) 
     5xGig-E (6.25 Gbps) 
     OBSAI RP3-01 (6.144 Gbps) 
     6xGig-E (7.50 Gbps) 
     Fibre Channel 8GFC (8.5 Gbps) 
     SONET OC-192 (9.952 Gbps) 
     Fibre Channel 10GFC Serial (10.52 Gbps) 
     The High Data Rate Wireless Communications Radio described in this patent will support all of these protocols and a variety of others, up to a maximum data rate of about 13 Gbps. In preferred embodiments operating at data rates in the range of about 3.5 Gbps the occupied transmit bandwidth should be between 1.0 GHz and 5 GHz. For the higher data rates the transmit bandwidth will preferably be in a range closer to the 5 GHz limit 
     Many if not most of the components to the transmitter and the receiver could be fabricated on a single integrated circuit chip to greatly reduce the cost of individual radio units. Other features of millimeter wave and application of millimeter wave radios that have been described in the patents listed in the background section could be applied to the radios described above. 
     Therefore readers should determine the scope of the present invention by reference to the appended claims.