Abstract:
A power electronics device with an improved IGBT protection mechanism is provided. More specifically, systems and methods are provided for reducing the switching frequency of an inverter module based on the junction temperature variation of the IGBT.

Description:
BACKGROUND 
     The invention relates generally to the field of electrical power converters and inverters. More particularly, the invention relates to techniques for preventing failure of motor drive circuitry due to overheating. 
     Power inverters and converters typically employ power modules to create a desired output current waveform, which is used to power various devices, such as motors and other equipment. The frequency and amplitude of the output current waveform may effect the operation of the device such as by changing the speed or torque of a motor, for example. Some power modules create the desired output current waveform through pulse width modulation, wherein power semiconductor switches such as insulated gate bipolar transistors (IGBTs) are caused to switch rapidly on and off in a particular sequence so as to create an approximately sinusoidal output current waveform. Furthermore, high IGBT switching speeds tend to produce a smoother, more ideal sinusoidal waveform, which may be desirable in some applications. For example, in heating, ventilating, and air conditioning systems a smoother sinusoidal waveform will reduce system noise and vibrations. 
     Higher IGBT switching speeds may tend, however, to increase the junction temperature of the IGBTs, which may result in more mechanical stress and increased rates of IGBT failure over time. Attempts have been made to reduce IGBT failure by limiting the maximum absolute IGBT junction temperatures. However, these techniques have failed to account for the increased stresses that tend to occur under start-up conditions or low-speed conditions, wherein the IGBTs tend to experience high current at low output frequency. 
     It may be advantageous, therefore, to provide a system and method of reducing IGBT thermal stress that is particularly effective under start-up conditions and low-speed, high-current conditions. Specifically, it may be advantageous to provide a method of reducing temperature variations of the IGBT junction, i.e. the semiconductor chip itself, and the case, i.e. the package in which the semiconductor chip is contained. 
     BRIEF DESCRIPTION 
     The present invention relates generally to an IGBT protection mechanism configuration designed to address such needs. Embodiments include systems and methods of reducing the switching frequency of an inverter module to avoid high junction temperature variation. Embodiments also include methods of estimating the expected junction temperature variation. 
    
    
     
       DRAWINGS 
       These and other features, aspects, and advantages of the present invention will become better understood when the following detailed description is read with reference to the accompanying drawings in which like characters represent like parts throughout the drawings, wherein: 
         FIG. 1  is a block diagram of a motor drive system employing circuitry for preventing high junction temperature variation in accordance with embodiments of the present invention; 
         FIG. 2  is a simplified circuit diagram illustrating an inverter module of the motor drive system of  FIG. 1 ; 
         FIG. 3  is a side view of an IGBT of the inverter module shown in  FIG. 2 , illustrating the failure modes associated therewith; 
         FIG. 4  depicts a thermal network model of the inverter module shown in  FIG. 2 ; 
         FIG. 5  is a graph illustrating the junction temperature variation in relation to the output current; 
         FIG. 6  is a graph illustrating the maximum junction temperature variation as a function of the switching frequency and the output frequency; and 
         FIG. 7  is a flowchart depicting one method of controlling the switching frequency of an inverter module. 
     
    
    
     DETAILED DESCRIPTION 
     Embodiments of the present invention relate to reducing the mechanical stress on IGBTs due to large temperature variations of the junction. Large junction temperature variations may contribute to particularly high levels of mechanical stress, because the different expansion rates of the various materials inside the IGBT package may lead to wire crack growth in wire bonds and similar contacts. Therefore, reducing junction temperature variations may result in a longer lasting inverter module. In embodiments of the present invention, the junction temperature variation is controlled by controlling the switching frequency. Because the highest junction temperature variations tend to occur during start-up or low-speed, high-current conditions, the switching frequency may be reduced for only a short time during start-up, after which the switching frequency may be increased to provide a smoother sinusoidal waveform. 
       FIG. 1  illustrates an exemplary motor control system  10  employing circuitry for preventing extreme junction temperature variation under variable operating conditions. A three-phase power supply  12  provides a three-phase voltage waveform at a constant frequency to a rectifier  14 , and may be derived from a generator or from an external power grid. Rectifier  14  performs full wave rectification of the three phase voltage waveform, outputting a direct current (DC) voltage to an inverter module  16 . 
     Inverter module  16  accepts the positive and negative lines of DC voltage from the rectifier circuitry  14  and outputs a discretized three phase waveform at a desired frequency, independent of the frequency of three-phase power supply  12 . Driver circuitry  18  provides inverter module  16  with appropriate signals, enabling inverter module  16  to output the waveform. The resulting three-phase waveform may thereafter drive a load, such as a motor  20 . 
     Control circuitry  22  may receive commands from remote control circuitry  24 , using such commands to enable driver circuitry  18  to properly control inverter module  16 . In some embodiments, the motor control system may include one or more sensors  26  for detecting operating temperatures, voltages, currents, etc. With feedback data from sensors  26 , control circuitry  22  may keep detailed track of the various conditions under which inverter module  16  may be operating. The feedback data may further allow control circuitry  22  to determine when inverter module  16  may be approaching a high temperature, allowing the control circuitry to implement preventative measures. 
     Referring to  FIG. 2 , an inverter module  16  may include a plurality of IGBTs  28  and power diodes  30 . The IGBTs  28  and power diodes  30  are joined to positive or negative DC lines (as appropriate) and output lines a, b, or c with bond wires  32 . As IGBTs  28  are rapidly switched on and off to produce a discretized three-phase output current waveform at the output  34 , strain is placed on bond wires  32  as a result of deformation resulting from stresses of thermal cycling. 
       FIG. 3  illustrates a side view of a portion of the inverter module  16  exhibiting signs of bond wire failure due to cumulative deformation resulting from heat stress. The inverter module  16  may include a direct bond copper (DBC) substrate  38 , which may include a ceramic base  40 , a copper layer  42 , and copper contacts  44  and  46 . Above the DBC substrate  38 , copper  44  is joined by solder  48  to the silicon IGBT  28 . Bond wire  52  joins the IGBT  28  to the copper contact  46 . High junction temperatures may tend to cause the DBC substrate  38  and the bond wire  52  to heat and expand unevenly, creating tension on the bond wire  52 , particularly at the solder connections. As the temperature difference between the IGBT  28  and the DBC substrate  38  increases, the tension on the bond wire  52  also increases due to the different rates of thermal expansion of the bond wire  52  and the DBC substrate  38 . Therefore, the junction temperature variation of the IGBT  28 , may have a pronounced effect on the life of the inverter module. As will be discussed further below, the junction temperature variation may tend to be greatest during startup or low-speed, high-current conditions. After numerous power cycles are applied to the IGBT  28 , deformation of the bond wire  52  may tend to cause heel cracks  54  in bond wire  52 . Additionally, the bond wire  52  may begin to separate from silicon chip  46  or metal plate  50  due to lift-off  56 . When a heel crack  54  or lift-off  56  severs bond wire  52  completely, the IGBT  28  may become inoperable. 
       FIG. 4  illustrates a thermal network model  58  illustrating the junction-to-case thermal impedance Z jc    60 . The junction-to-case thermal impedance Z jc    60  includes four thermal impedances in series, each corresponding with one of the physical layers shown in  FIG. 3 . Each thermal impedance includes a thermal resistance and a thermal capacitance joined in parallel. Thermal resistances include R 1    62 , R 2    64 , R 3    66 , and R 4    68 , and thermal capacitances include C 1    70 , C 2    72 , C 3    74 , and C 4    76 , values for which may generally be obtained from a datasheet provided by the manufacturer of inverter module  16 . The thermal network may be used to estimate the temperature difference between a junction of the solid state IGBT and the case to which the IGBT is mounted, as will be explained further below. It will be appreciated that the thermal network  58  may be different for different inverter modules, and may include more or fewer thermal impedance elements. 
       FIG. 5  depicts a graph  78  showing the junction temperature variation of a typical inverter module  16  with respect to time, overlaid with the current output of the inverter module  16 . The graph  78  includes a horizontal axis  80 , which represents time, and two vertical axes  82  and  84 . The right side vertical axis  82  represents current and the left vertical axis  84  represents temperature. Trace  86  represents the current output of one phase of the inverter module. It should be noted that because the current output varies about a zero point, two IGBTs  28  may be utilized, one for the positive polarity and one for the negative polarity. Trace  88  represents the junction temperature of one of the IGBTs  28  producing the current output depicted by trace  86 . It should also be noted that because the IGBT  28  represented is in the off state during the negative polarity current output, the IGBT  28  continues to cool during this time period, as shown by trace  88 . It will be appreciated that the current generated on the diodes  30  will also result in similar temperature characteristics. 
     As can be seen in the graph  78 , the junction temperature reaches a peak  90  during each cycle of the output current and reaches a valley  91  after each half cycle. The junction temperature variation is defined as the difference between the peak  90  temperature and the valley  91  temperature. It will be appreciated that the thermal stresses discussed above may be greatest when the junction temperature reaches the peak  90 . Therefore, some embodiments may include estimating the peak IGBT  28  junction temperature for one cycle of the output current. Still other embodiments may include estimating the peak diode  30  temperature for one cycle of the output current. 
     To estimate the IGBT  28  junction temperature variation, the thermal network  58 , described above in relation to  FIG. 4 , may be factored by the peak junction temperature, as will be described below. Accordingly, embodiments of the present invention also include a method of estimating the peak junction temperature. In some embodiments, the estimated peak junction temperature may be based on the estimated power losses of the IGBTs  28 . Furthermore, the estimated power losses of the IGBT  28  may be based on estimated operating conditions of the IGBTs  28 . For example, peak IGBT  28  junction temperature estimates may be based on estimated conduction losses and switching losses as calculated according to the following equations: 
                         P   c     ⁡     (     f   ,     I     R   ⁢           ⁢   MS         )       =         (       1     2   ·   π       +         M   ⁡     (   f   )       ·   Pf     8       )     ·     V   t     ·     2     ·     I     R   ⁢           ⁢   MS         +       (       1   8     +         M   ⁡     (   f   )       ·   PF       3   ·   π         )     ·     R   t     ·   2   ·     I     R   ⁢           ⁢   MS     2           ;           (   1   )                       ⁢           P   s     ⁡     (       f   s     ,     I     R   ⁢           ⁢   MS         )       =       1   π     ·     f   s     ·     E   onoff     ·     (         2     ·     I     R   ⁢           ⁢   MS           I   nom       )     ·     (       V     D   ⁢           ⁢   C         V   nom       )         ;             (   2   )                       ⁢         P   ⁡     (     f   ,     f   s     ,     I     R   ⁢           ⁢   MS         )       =         P   c     ⁡     (     f   ,     I     R   ⁢           ⁢   MS         )       +       P   s     ⁡     (       f   s     ,     I     R   ⁢           ⁢   MS         )           ;             (   3   )               
wherein P c  equals the estimated conduction power loss as a function of the fundamental frequency, f, and the output RMS current of the drive, I RMS ; P s  equals the estimated switching power losses as a function of the switching frequency, f s , and the output RMS current of the drive, I RMS ; and P(f, f s , I RMS ) equals the total estimated power losses of the IGBT  28 . In equation (1), M(f) represents the modulation index and PF represents the power factor of a load driven by the inverter module  16 . In equation (1), V t  represents the approximate IGBT  28  conduction voltage at small or near zero forward current and R t  represents the approximate slope resistance. Both V t  and R t  may be derived from a manufacturer datasheet. In equation (2), E onoff  represents the total energy required to switch the IGBT  28  on and off at a rated voltage V nom  (half of the IGBT rated voltage) and current I nom  (rated IGBT module current) of the IGBT  28 . All three of E onoff , V nom , and I nom  may be obtained from manufacturer data sheets. I RMS  and V DC  represent the estimated operating current and bus voltage of the IGBT  28 . Therefore, both I RMS /I nom  and V DC /V nom  act as scaling factors applied to the switching loss value, E onoff , may obtained from the manufacturing data sheet. The total power loss, P, may then be used to calculate the junction temperature variation using the thermal network  16 , as described below.
 
     In some embodiments, the calculation of the junction temperature variation, (ΔT j ), may be simplified by assuming that the temperature variation of the case is negligible. As such, a “boost factor” (BF(f)) may be first calculated, based on the thermal network  58  according to the following equation: 
                       BF   ⁡     (   f   )       =     1   +       ∑     i   =   1     4     ⁢         R   i       R   jc       ·       π   -   1         1   +       (     2   ⁢     π   ·   f   ·     τ   i         )     2                   ;           (   4   )               
Wherein R i  and τ i  equal the thermal resistances and capacitances of the thermal network, as shown in  FIG. 4 , and R ic  equals the overall thermal resistance between the junction and the case. Furthermore, an interim value, BF_ΔT j , may be calculated from the boost factor, according to the following equations:
 
 BF   —   ΔT   j ( f )=1.85·( BF ( f )−1) if  BF ( f )&lt;2  (5);
 
 BF   —   ΔT   j ( f )= BF ( f ) if  BF ( f )&gt;2  (6).
 
     Having obtained the estimated power losses and the boost factor, the estimated junction temperature variation, ΔT j , may then be approximated according to the following formula:
 
 ΔT   j ( f, f   s   , I   rms )= PI ( f, f   s   , I   rms )· BF   —   ΔT   j ( f )· R   j   (15).
 
Wherein, ΔTj represents the junction temperature variation after one output cycle of the inverter module.
 
     It will be appreciated that variations of the above formulas may be made while still falling within the scope of the present invention. Additionally, in some embodiments one or more of the variables, such as I RMS , E onoff  or V DC  for example, may be measured. Alternatively, these variables may also be estimated based on average known operating conditions of typical inverter modules or a particular inverter module. Additionally, in some embodiments, the diode  30  junction temperature variation may estimated rather than the IGBT junction temperature variation. 
     Turning now to  FIG. 6 , a graph  92  showing the junction temperature variation, ΔTj, of a typical inverter module is depicted. Specifically, the graph  92  depicts ΔTj versus the fundamental frequency, i.e. the output frequency of the inverter module, at IGBT switching frequencies of 4 kHz (trace  98 ), 8 kHz (trace  100 ), and 12 kHz (trace  102 ). 
     As can be seen in graph  92 , ΔTj may tend to be greater when the inverter module is operating at a low fundamental frequency and greatest for the DC condition. This may be due to the fact that current will tend to be concentrated on an individual IGBT  28  for a longer time period when the inverter is operating at lower output frequencies. Because typical motor drives may operate at lower speeds during startup, higher junction temperature variation may also be present during startup. 
     Furthermore, as can also be seen in the graph  92 , ΔTj may tend to be greater for higher switching frequencies. This may be due to the fact that most of the power losses in an IGBT  28  occur during the brief transitional period when the IGBT  28  is switching on or off. Therefore, higher switching frequencies tend to result in higher junction temperatures and higher junction temperature variations. 
     It will also be appreciated that IGBTs  28  generally conduct higher levels of current at startup due, in part, to the lower frequency and low impedance of the motor windings under this condition, wherein the motor windings are magnetized. All of the above factors may contribute to higher levels of inverter module wear during startup conditions, which, as discussed above, may eventually lead to failure. Therefore, in order to reduce excessive inverter module wear at startup, embodiments of the present invention include temporarily reducing the switching frequency of the inverter module  16 , thereby reducing the junction temperature variation and thermal stress on the IGBTs  28 . 
       FIG. 7  depicts a flow chart of a process  104  for reducing the switching frequency (depicted in  FIG. 7  as “f s ”) in accordance with embodiments of the present invention. In some embodiments, the process  104  may be implemented in the control circuitry  22 , discussed in relation to  FIG. 1 . First, at step  106 , the power losses of the IGBTs  28  may be estimated using equation (3) as described above. In some embodiments, the power losses estimated at step  106  may be based on known or estimated operating values and conditions. In other embodiments, one or more values, such as RMS current for example, may be based on measurements, such as measurements performed by the sensor(s)  26  (see  FIG. 1 ) and communicated back to the control circuitry  22 . 
     Next, at step  108 , the power losses may be used to calculate an estimated junction temperature variation, ΔT j . Then, at steps  110 - 116 , the estimated junction temperature variation is used to determine the switching frequency. Embodiments of the present invention may toggle the switching frequency between two alternate values: a command frequency (depicted as “f cmd ”) and a low frequency (depicted as “f low ”). The command frequency is the switching frequency at which the inverter module is intended to be operated under most conditions. For example, the command frequency may be a relatively high switching frequency, on the order of four thousand to twelve thousand Hertz, used to create a smooth sinusoidal waveform, which may minimize noise in some systems. The low frequency is the temporary switching frequency that is used to minimize IGBT stress. For example, in some embodiments, the low frequency may be approximately two thousand Hertz. Both the command frequency and the low frequency may be programmed into the driver circuitry  18  or the control circuitry  22  and may be specified by the user, such as through the remote control circuitry  24  (see  FIG. 1 ). 
     At step  110  it is determined whether the switching frequency (f s ) equals the low frequency (f low ). If the switching frequency equals the low frequency at step  110 , then process  104  branches to steps  116  and  118 , wherein the switching frequency may be increased to the command frequency (f cmd ). Specifically, if ΔTj is below a specified temperature at step  116 , in this case fifty-five degrees Celsius, then the process  104  proceeds to step  118 , at which point the switching frequency is set to the command frequency. However, if, at step  110 , the switching frequency does not equal the low frequency then process  104  branches to steps  112 - 114 , wherein the switching frequency may be reduced to the low frequency. Specifically, if ΔTj is above a specified temperature at step  112 , in this case sixty degrees Celsius, then process  104  proceeds to step  114 , at which point the switching frequency is set to the low frequency. The process  104  may then repeat, starting at step  106 . 
     According to process  104 , if the junction temperature variation rises above sixty degrees Celsius, the switching frequency will be reduced. Thereafter, if the junction temperature variation drops below fifty-five degrees Celsius then the switching frequency will increase back to the command frequency. Referring to  FIG. 6 , it will be appreciated that the result of the process  104  described above is that the switching frequency may be low only for a short time after start-up. 
     It will also be appreciated that variations of the above process may be made while still falling within the scope of the present invention. For example, some embodiments may include varying the switching frequency of the inverter module between three or more frequency values, depending on the estimated junction temperature variation. For another example, some embodiments may include predetermining the expected junction temperature variation for given operating conditions or a given inverter module. Furthermore, the switching frequency may be initially set to the low frequency at startup and increased to the command frequency when the fundamental frequency of the inverter module  16  is above a specified frequency chosen based on the expected junction temperature variation for that fundamental frequency. Furthermore, in other embodiments, the diode  30  junction temperature variation may be used to reduce the switching frequency instead of the IGBT junction temperature variation. 
     While only certain features of the invention have been illustrated and described herein, many modifications and changes will occur to those skilled in the art. It is, therefore, to be understood that the appended claims are intended to cover all such modifications and changes as fall within the true spirit of the invention.