Abstract:
A circuit is designed with a measurement circuit ( 746 ) coupled to receive an input signal from at least one of a first antenna and a second antenna of a transmitter. The measurement circuit produces an output signal corresponding to a magnitude of the input signal. A control circuit ( 726 ) is coupled to receive the output signal, a first reference signal (.eta..sub.1) and a second reference signal (.eta..sub.2). The control circuit is arranged to produce a control signal in response to a comparison of the output signal, the first reference signal and the second reference signal.

Description:
This application is a divisional of application Ser. No. 10/808,621, filed Mar. 24, 2004, now U.S. Pat. No. 7,555,068, issued Jun. 30, 2009; 
     Which is a divisional of application Ser. No. 09/373,855, filed Aug. 13, 1999, now U.S. Pat. No. 6,728,302, granted Apr. 27, 2004. Which claims priority under 35 U.S.C. .sctn. 119(e)(1) of provisional application No. 60/119,732, filed Feb. 12, 1999 and provisional application No. 60/120,609, filed Feb. 18, 1999. 
    
    
     FIELD OF THE INVENTION 
     This invention relates to wideband code division multiple access (WCDMA) for a communication system and more particularly to space-time transmit diversity (STTD) detection for WCDMA signals. 
     BACKGROUND OF THE INVENTION 
     Present code division multiple access (CDMA) systems are characterized by simultaneous transmission of different data signals over a common channel by assigning each signal a unique code. This unique code is matched with a code of a selected receiver to determine the proper recipient of a data signal. These different data signals arrive at the receiver via multiple paths due to ground clutter and unpredictable signal reflection. Additive effects of these multiple data signals at the receiver may result in significant fading or variation in received signal strength. In general, this fading due to multiple data paths may be diminished by spreading the transmitted energy over a wide bandwidth. This wide bandwidth results in greatly reduced fading compared to narrow band transmission modes such as frequency division multiple access (FDMA) or time division multiple access (TDMA). 
     Previous studies have shown that multiple transmit antennas may improve reception by increasing transmit diversity for narrow band communication systems. In their paper  New Detection Schemes for Transmit Diversity with no Channel Estimation , Tarokh et al. describe such a transmit diversity scheme for a TDMA system. The same concept is described in  A Simple Transmitter Diversity Technique for Wireless Communications  by Alamouti. Tarokh et al. and Alamouti, however, fail to teach such a transmit diversity scheme for a WCDMA communication system. 
     New standards are continually emerging for transmit diversity of next generation wideband code division multiple access (WCDMA) communication systems as described in Provisional U.S. Patent Application No. 60/116,268, filed Jan. 19, 1999, and incorporated herein by reference. These WCDMA systems are coherent communications systems with pilot symbol assisted channel estimation schemes. These pilot symbols are transmitted as quadrature phase shift keyed (QPSK) known data in predetermined time frames to any receivers within range. The frames may propagate in a discontinuous transmission (DTX) mode. For voice traffic, transmission of user data occurs when the user speaks, but no data symbol transmission occurs when the user is silent. Similarly for packet data, the user data may be transmitted only when packets are ready to be sent. The frames are subdivided into sixteen equal time slots of 0.625 milliseconds each. Each time slot is further subdivided into equal symbol times. At a data rate of 32 KSPS, for example, each time slot includes twenty symbol times. Each frame includes pilot symbols as well as other control symbols such as transmit power control (TPC) symbols and rate information (RI) symbols. These control symbols include multiple bits otherwise known as chips to distinguish them from data bits. The chip transmission time (T C ), therefore, is equal to the symbol time rate (T) divided by the number of chips in the symbol (N). 
     A mobile unit must initially receive and synchronize with data frames transmitted by one or more remote base stations. Each base station continually transmits broadcast channel (BCH) data over the primary common control physical channel (PCCPCH) to identify itself to mobile units within the cell. Referring to  FIG. 1 , there is a simplified block diagram of a typical diversity transmitter of the prior art. The transmitter simultaneously transmits primary and secondary synchronization codes on respective primary (P-SCH)  150  and secondary (S-SCH)  160  channels to uniquely identify each base station signal received by the mobile unit. Circuits  156  and  166  modulate the gain of these synchronization codes in response to respective gain factors GP-SCH on lead  154  and GP-SCH on lead  164 . Circuit  170  adds the synchronization codes and applies them to time switch (TSW)  174  via lead  172 . Time switch  174  selectively applies the synchronization codes to switches SW 0   134  and SW 1   136  in response to the control signal at lead  140  as indicated by inset  190 . These P-SCH and S-SCH codes are transmitted as symbol  300  ( FIG. 3 ) in time slot  1 . 
     Broadcast channel data (BCH) for the PCCPCH are applied to channel encoder  108  via lead  106  ( FIG. 1 ). Interleaver circuit  110  applies the BCH data to space-time transmit diversity (STTD) encoder circuit  112 . The STTD encoder produces encoded output data at lead  114  for the transmit antenna (Ant  1 ) and at lead  116  for the diversity antenna (Ant  2 ). Multiplex circuit  118  produces this STTD encoded BCH data on leads  120  and  122  at a time corresponding to data symbols  302  of time slot  1  ( FIG. 3 ). The BCH data are modulated by spreading and scrambling codes on lead  124  and applied to switches SW 0   134  and SW 1   136 . These switches SW 0  and SW 1  selectively multiplex SCH data with BCH data and pilot symbols in response to a control signal on lead  138  as shown at inset  190 . The BCH data at lead  180  are then applied to the transmit antenna (Ant  1 ), and the data at lead  182  is applied to the diversity antenna (Ant  2 ). 
     Pilot symbol data for the PCCPCH are applied to lead  100 . Diversity circuit  102  generates an open loop transmit diversity (OTD) symbol pattern at lead  104  for the diversity antenna. This OTD pattern together with the pilot symbol pattern for the transmit antenna is shown at TABLE I for each of the sixteen time slots in a frame. By way of comparison, the STTD pilot symbol pattern for diversity antenna (Ant  2 ) transmission on the dedicated physical data channel (DPDCH) is also shown. The pilot symbols at leads  100  and  102  are applied to multiplex circuit  118 . Multiplex circuit  118  selectively applies the pilot symbols at leads  100  and  102  to leads  120  and  122 , respectively, at a time corresponding to pilot symbols  304  of time slot  1  ( FIG. 3 ). Thus, multiplex circuit  118  multiplexes STTD encoded data symbols  302  with OTD encoded pilot symbols  304 . The pilot symbols at leads  120  and  122  are then modulated with spreading and scrambling code. These modulated pilot symbols at leads  130  and  132  are further multiplexed with SCH data by switches  134  and  136 , respectively, in response to the control signal at lead  138  as shown at inset  190 . The resulting pilot symbols are applied to transmit and diversity antennas via leads  180  and  182 , respectively. 
     
       
         
               
               
               
               
             
               
               
               
               
               
               
               
               
               
               
               
               
               
             
               
               
               
               
               
               
               
               
               
               
               
               
               
             
           
               
                   
                 TABLE 1 
               
             
             
               
                   
                   
               
               
                   
                 TRANSMIT ANTENNA 
                 STTD ANT 2 
                 OTD ANT 2 
               
             
          
           
               
                 SLOT 
                 B 1   
                 S 1   
                 B 2   
                 S 2   
                 B 1   
                 −S 2 * 
                 −B 2   
                 S 1 * 
                 B 1   
                 S 1   
                 −B 2   
                 −S 2   
               
               
                   
               
             
          
           
               
                 1 
                 11 
                 11 
                 11 
                 11 
                 11 
                 01 
                 00 
                 10 
                 11 
                 11 
                 00 
                 00 
               
               
                 2 
                 11 
                 11 
                 11 
                 01 
                 11 
                 11 
                 00 
                 10 
                 11 
                 11 
                 00 
                 10 
               
               
                 3 
                 11 
                 01 
                 11 
                 01 
                 11 
                 11 
                 00 
                 00 
                 11 
                 01 
                 00 
                 10 
               
               
                 4 
                 11 
                 10 
                 11 
                 01 
                 11 
                 11 
                 00 
                 11 
                 11 
                 10 
                 00 
                 10 
               
               
                 5 
                 11 
                 10 
                 11 
                 11 
                 11 
                 01 
                 00 
                 11 
                 11 
                 10 
                 00 
                 00 
               
               
                 6 
                 11 
                 10 
                 11 
                 11 
                 11 
                 01 
                 00 
                 11 
                 11 
                 10 
                 00 
                 00 
               
               
                 7 
                 11 
                 01 
                 11 
                 00 
                 11 
                 10 
                 00 
                 00 
                 11 
                 01 
                 00 
                 11 
               
               
                 8 
                 11 
                 10 
                 11 
                 01 
                 11 
                 11 
                 00 
                 11 
                 11 
                 10 
                 00 
                 10 
               
               
                 9 
                 11 
                 11 
                 11 
                 00 
                 11 
                 10 
                 00 
                 10 
                 11 
                 11 
                 00 
                 11 
               
               
                 10 
                 11 
                 01 
                 11 
                 01 
                 11 
                 11 
                 00 
                 00 
                 11 
                 01 
                 00 
                 10 
               
               
                 11 
                 11 
                 11 
                 11 
                 10 
                 11 
                 00 
                 00 
                 10 
                 11 
                 11 
                 00 
                 01 
               
               
                 12 
                 11 
                 01 
                 11 
                 01 
                 11 
                 11 
                 00 
                 00 
                 11 
                 01 
                 00 
                 10 
               
               
                 13 
                 11 
                 00 
                 11 
                 01 
                 11 
                 11 
                 00 
                 01 
                 11 
                 00 
                 00 
                 10 
               
               
                 14 
                 11 
                 10 
                 11 
                 00 
                 11 
                 10 
                 00 
                 11 
                 11 
                 10 
                 00 
                 11 
               
               
                 15 
                 11 
                 01 
                 11 
                 00 
                 11 
                 10 
                 00 
                 00 
                 11 
                 01 
                 00 
                 11 
               
               
                 16 
                 11 
                 00 
                 11 
                 00 
                 11 
                 10 
                 00 
                 01 
                 11 
                 00 
                 00 
                 11 
               
               
                   
               
             
          
         
       
     
     Turning now to  FIG. 2 , there is a block diagram showing signal flow in an OTD encoder  102  of the prior art for pilot symbol encoding of the transmitter of  FIG. 1 . The pilot symbols are predetermined control signals that may be used for channel estimation and other functions as will be described in detail. The OTD encoder  102  receives pilot symbols B 1 , S 1 , B 2  and S 2  at symbol times T-4T, respectively, on lead  100 . These pilot symbols are applied to the transmit antenna (Ant  1 ) via multiplex circuit  118  and switch SW 0   134  as previously described. The OTD encoder  102  simultaneously produces pilot symbols B 1 , S 1 , −B 2  and −S 2  at symbol times T-4T, respectively, at lead  104  for the OTD diversity antenna (Ant  2 ). The pilot symbol pattern for the transmit and OTD diversity antennas is shown at TABLE I for the sixteen time slots of a frame. Each symbol includes two bits representing a real and imaginary component. An asterisk indicates a complex conjugate operation or sign change of the imaginary part of the symbol. Pilot symbol values for the first time slot for the transmit antenna at lead  104 , therefore, are 11, 11, 11 and 11. Corresponding pilot symbols for the second antenna at lead  104  are 11, 11, 00 and 00. 
     The bit signals r j (i+τ j ) of these symbols are transmitted serially along respective paths  208  and  210 . Each bit signal of a respective symbol is subsequently received at a remote mobile antenna  212  after a transmit time τ corresponding to the j th  path. The signals propagate to a despreader circuit ( FIG. 6 ) where they are summed over each respective symbol time to produce input signals R j   1 , R j   2 , R j   3  and R j   4  corresponding to the four pilot symbol time slots and the j th  of L multiple signal paths. 
     The input signals corresponding to the pilot symbols for each time slot are given in equations [1-4]. Noise terms are omitted for simplicity. Received signals R j   1 , R j   2 , R j   3  and R j   4  are produced by respective pilot symbols B 1 , S 1 , B 2  and S 2 . Average channel estimates {circumflex over (α)} j   1  and {circumflex over (α)} j   2  over the four pilot symbols for each antenna are obtained from a product of each received signal and a complex conjugate of its respective pilot symbol as in equations [5] and [6].
 
 R   j   1 =(α j   1 +α j   2 ) B   1   [1]
 
 R   j   2 =(α j   1 +α j   2 ) S   1   [2]
 
 R   j   3 =(α j   1 −α j   2 ) B   2   [3]
 
 R   j   4 =(α j   1 −α j   2 ) S   2   [4]
 
{circumflex over (α)} j   1 =( B   1   *R   j   1   +S   1   *R   j   2   +B   2   *R   j   3   +S   2   *R   j   4 )/4  [5]
 
{circumflex over (α)} j   2 =( B   1   *R   j   1   +S   1   *R   j   2   −B   2   *R   j   3   −S   2   *R   j   4 )/4  [6]
 
     Referring now to  FIG. 4 , there is a simplified diagram of a mobile communication system of the prior art. The mobile communication system includes an antenna  400  for transmitting and receiving external signals. The diplexer  402  controls the transmit and receive function of the antenna. Multiple fingers of rake combiner circuit  404  combine received signals from multiple paths. Symbols from the rake combiner circuit  404 , including pilot symbol signals, are applied to a bit error rate (BER) circuit  410  and to a Viterbi decoder  406 . Decoded symbols from the Viterbi decoder are applied to a frame error rate (FER) circuit  408 . Averaging circuit  412  produces one of a FER and BER. This selected error rate is compared to a corresponding target error rate from reference circuit  414  by comparator circuit  416 . The compared result is applied to bias circuit  420  via circuit  418  for generating a signal-to-interference ratio (SIR) reference signal on lead  424 . 
     Pilot symbols from the rake combiner  404  are applied to the SIR measurement circuit  432 . The SIR measurement circuit produces a received signal strength indicator (RSSI) estimate from an average of received pilot symbols. The SIR measurement circuit also produces an interference signal strength indicator (ISSI) estimate from an average of interference signals from base stations and other mobile systems over many time slots. The SIR measurement circuit produces an SIR estimate from a ratio of the RSSI signal to the ISSI signal. This SIR estimate is compared with a target SIR by circuit  426 . This comparison result is applied to TPC command circuit  430  via circuit  428 . The TPC command circuit  430  sets a TPC symbol control signal that is transmitted to a remote base station. This TPC symbol instructs the base station to either increase or decrease transmit power by 1 dB for subsequent transmission. 
     Turning now to  FIG. 5 , there is a diagram showing a weighted multi-slot averaging (WMSA) circuit  732  of the prior art for channel estimation. In operation, a signal buffer circuit  706  ( FIG. 7 ) receives individual frames of data having a predetermined time period of 10 milliseconds. Each frame of the PCCPCH is subdivided into sixteen equal time slots of 0.625 milliseconds each. Each time slot, for example time slot  528 , includes a respective set of pilot symbols  520  and data symbols  529 . The WMSA circuit ( FIG. 5 ) samples pilot symbols from preferably 6 time slots for a Doppler frequency of less than 80 Hz and from preferably 4 time slots for a Doppler frequency of 80 Hz or more. These sampled pilot symbols are multiplied by respective weighting coefficients α 1 , through α N  and combined by circuit  526  to produce a channel estimate. This channel estimate is used to correct the phase of received data symbols in time slot  527  estimate for a respective transmit antenna. 
     Referring now to  FIG. 6 , there is a despreader circuit of the prior art. Received signals from mobile antenna  212  propagate to the despreader circuit where they are summed over each respective symbol time to produce output signals R j   1  and R j   2  corresponding to the j th  of L multiple signal paths as previously described. The despreader circuit receives the i th  of N chip signals per symbol together with noise along the j th  of L multiple signal paths at a time τ j  after transmission. Both here and in the following text, noise terms are omitted for simplicity. This received signal r j (i+τ j ) at lead  600  is multiplied by a channel orthogonal code signal C m (i+τ j ) at lead  604  that is unique to the receiver. Each chip signal is summed over a respective symbol time by circuit  608  and produced as first and second output signals R j   1  and R j   2  on leads  612  and  614  as in equations [1-2], respectively. Delay circuit  610  provides a one-symbol delay T so that the output signals are produced simultaneously. 
     This arrangement advantageously provides additional gain at the mobile communication system by multiple path transmit antenna diversity from a remote base station. The mobile unit, however, must be compatible with base stations having a single transmit antenna as well as base stations having transmit antenna diversity. A problem arises, therefore, when the mobile communication system is initially powered up or when it passes from one cell to another cell. The mobile unit must not only determine which of several base signals offers a preferable signal strength. It must also determine whether the base station offers transmit antenna diversity. If the mobile unit incorrectly decodes a received signal and assumes no transmit diversity, it loses the improved gain of transmit diversity. Alternatively, if the mobile unit incorrectly decodes a received signal and assumes transmit diversity, multiple fingers of the rake combiner circuit  404  contribute noise to the received signal. 
     SUMMARY OF THE INVENTION 
     The foregoing problems are resolved by a circuit designed with a measurement circuit coupled to receive an input signal from at least one of a first antenna and a second antenna of a transmitter. The measurement circuit produces an output signal corresponding to a magnitude of the input signal. A control circuit is coupled to receive the output signal, a first reference signal and a second reference signal. The control circuit is arranged to produce a control signal in response to a comparison of the output signal, the first reference signal and the second reference signal. 
     The present invention detects a diversity transmit antenna. A control signal modifies receiver signal processing to correspond to the presence or absence of the diversity transmit antenna. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       A more complete understanding of the invention may be gained by reading the subsequent detailed description with reference to the drawings wherein: 
         FIG. 1  is a simplified block diagram of a typical transmitter of the prior art using OTD encoded pilot symbols and STTD encoded data symbols for the PCCPCH; 
         FIG. 2  is a block diagram showing signal flow of pilot symbol encoding in the OTD encoder of the transmitter of  FIG. 1 ; 
         FIG. 3  is a diagram of pilot, data and search channel symbols of a PCCPCH time slot; 
         FIG. 4  is a simplified block diagram of a receiver of the prior art; 
         FIG. 5  is a block diagram showing weighted multi-slot averaging (WMSA) of the prior art; 
         FIG. 6  is a schematic diagram of a despreader circuit of the prior art. 
         FIG. 7A  is a block diagram of a transmit diversity detection circuit of the present invention; 
         FIG. 7B  is a block diagram of another embodiment of a transmit diversity detection circuit of the present invention; 
         FIG. 7C  is a block diagram of the measurement circuit  746  of  FIG. 7A ; 
         FIG. 8A  is a simulation showing cumulative probability of detecting the presence of transmit diversity as a function of time for the embodiment of  FIG. 7A ; 
         FIG. 8B  is a simulation showing cumulative probability of not detecting transmit diversity when present for the embodiment of  FIG. 7A ; 
         FIG. 9A  is a simulation showing cumulative probability of detecting the absence of transmit diversity as a function of time for the embodiment of  FIG. 7A ; 
         FIG. 9B  is a simulation showing cumulative probability of detecting transmit diversity when absent; 
         FIG. 10A  is a simulation comparing normal and STTD decoding of single antenna transmission for a Doppler frequency of 5 Hz with weighted multi-slot averaging (WMSA); and 
         FIG. 10B  is a simulation comparing normal and STTD decoding of single antenna transmission for a Doppler frequency of 200 Hz with weighted multi-slot averaging (WMSA). 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Referring now to  FIG. 7A , there is a first embodiment of a mobile unit of the present invention configured for blind detection of transmit diversity. This blind detection scheme includes a new implementation of an algorithm disclosed by A. Wald,  Sequential Analysis  (1947). Mobile antenna  212  receives multipath signals transmitted by base station antennas at leads  180  and  182  ( FIG. 1 ), respectively. Diplexer circuit  702  couples these received multipath signals to lead  704  during receive mode operation. Doppler frequency estimator circuit  740  is described in detail in copending U.S. patent application Ser. No. 09/224,632, filed Dec. 31, 1998, and incorporated herein by reference. Doppler frequency estimator circuit  740  receives the multipath signals on lead  704  and produces an output signal on lead  742  corresponding to the estimated Doppler frequency. Delay profile estimator circuit  720  also receives the multipath signals on lead  704 . Delay profile estimator circuit  720  includes a despreader circuit as in  FIG. 6  and a match filter circuit (not shown). The delay profile estimator circuit  720  determines which of the received multipath signals should be combined based on the strength of the matched filter output. 
     Operation of the measurement circuit  746  will now be explained in detail with reference to  FIG. 7C . The measurement circuit  746  receives pilot symbol data from received multipath signals on lead  704 . Channel estimate circuit  750  generates separate diversity signals X 1  and X 2 , corresponding to antennas on leads  180  and  182 , respectively. These separate diversity signals include pilot symbols from a series of time slots. Coherent averaging circuit  756  coherently averages the energy of all received pilot symbol data from the respective antennas at leads  180  and  182  from K time slots in response to the Doppler frequency estimator circuit output signal on lead  742  and produces signals {tilde over (X)} 1  and {tilde over (X)} 1  on respective leads  758  and  760 . The variable K is preferably the same number of time slots used by the WMSA circuit of  FIG. 5 . It is preferably equal to six time slots for Doppler frequencies below 80 Hz and preferably equal to four time slots for Doppler frequencies of 80 Hz or more. Non-coherent averaging circuit  762  then non-coherently averages the signals {tilde over (X)} 1  and {tilde over (X)} 1  over the respective multipaths and produces signals |{tilde over (X)} 1 | and |{tilde over (X)} 2 | at leads  764  and  766 , respectively, in response to the output signal on lead  744  from the delay profile estimator circuit. Ratio circuit  768  produces an output signal λ at lead  722  that is a ratio of the signals |{tilde over (X)} 1 | and |{tilde over (X)} 2 | from the primary antenna at lead  180  and the diversity antenna at lead  182 , respectively. 
     Comparator circuit  726  compares the output signal λ at lead  722  to the first reference signal η 1  and the second reference signal η 2  at leads  723  and  724 , respectively. These reference signals are programmed such that reference signal η 1  is greater than reference signal η 2 . When output signal λ is greater than reference signal η 1 , the comparator circuit produces a control signal on lead  728  indicating no transmit diversity. This control signal is applied to WMSA channel estimate circuit  732 . The WMSA channel estimate circuit sets the channel estimate α j   2  at lead  736  to zero, thereby eliminating any noise contribution to the received signal. Phase correction circuit  710  then applies the channel estimate α j   1  at lead  734  to the received signal at lead  708  from signal buffer  706 . The phase correction circuit applies a corrected received signal from the primary antenna at lead  180  to rake combiner circuit  712 . This rake combiner circuit then combines corrected multi-path signals from the primary antenna and applies the resulting combined signal to Viterbi decoder circuit  714 . The Viterbi decoder produces a received signal at lead  716 . 
     Alternatively, when output signal λ is less than reference signal η 2 , the ratio of signals from the primary and diversity antennas is near unity. The comparator circuit  726 , therefore, produces a control signal on lead  728  indicating transmit diversity. The control signal is also applied to WMSA channel estimate circuit  732 . The WMSA channel estimate circuit responsively produces channel estimate signals α j   1  and α j   2  at leads  734  and  736 , respectively. Phase correction circuit  710  then applies both channel estimates to the received signal at lead  708  from signal buffer  706 . The phase correction circuit then applies corrected signals from the primary antenna at lead  180  and the diversity antenna at lead  182  to rake combiner circuit  712 . This rake combiner circuit then combines corrected multi-path signals from both antennas and applies the resulting combined signal to Viterbi decoder circuit  714 . The Viterbi decoder produces a received signal at lead  716 . 
     When output signal λ is less than reference signal η 1  but greater η 2 , the ratio of signals is indeterminate and comparator circuit  726  does not change the control signal on lead  728 . Thus, WMSA channel estimate circuit continues to produce channel estimates corresponding to the previous state. Likewise, phase correction circuit  710 , rake combiner  712  and Viterbi decoder  714  continue in the same mode of operation until output signal λ exceeds the bounds of one of the reference signals, thereby indicating an unambiguous presence or absence of diversity. Furthermore, reference signal η 1  and η 2  preferably converge to a single value η over time. This sequential convergence assures sequential detection of diversity or non-diversity over time. 
     The simulation output of  FIG. 8A  shows cumulative probability of detecting the presence of transmit diversity as a function of time for the embodiment of  FIG. 7A . The simulation conditions include 40 traffic channels, each having a gain equal to the PCCPCH. Reference signals η 1  and η 2  converge to η linearly over 48 frames for Doppler rates of 5 Hz and 20 Hz and over 24 frames for a vehicular Doppler rate of 200 Hz. The simulation shows 99% cumulative probability of detection of a diversity antenna at 250 milliseconds, 145 milliseconds and 30 milliseconds for Doppler frequencies of 5 Hz, 20 Hz and 200 Hz, respectively. The simulation of  FIG. 8B  shows cumulative probability P m  of not detecting transmit diversity when present for the embodiment of  FIG. 7A . The simulated probabilities are 1.7×10 −3  and 1.2×10 −4  for pedestrian Doppler frequencies of 5 Hz and 20 Hz, respectively. No error occurred at a 200 Hz Doppler frequency. 
     The simulation output of  FIG. 9A  shows cumulative probability of detecting the absence of transmit diversity as a function of time for the embodiment of  FIG. 7A . Under the same simulation conditions as  FIG. 8 , the simulation shows 99% cumulative probability of detecting the absence of a diversity antenna at 170 milliseconds, 140 milliseconds and 55 milliseconds for Doppler frequencies of 5 Hz, 20 Hz and 200 Hz, respectively. The simulation of  FIG. 9B  shows cumulative probability P f  of detecting transmit diversity when not present for the embodiment of  FIG. 7A . The simulated probabilities are 6.5×10 −3 , 3.6×10 −3  and 6.1×10 −4  for Doppler frequencies of 5 Hz, 20 Hz and 200 Hz, respectively. No error occurred at a 200 Hz Doppler frequency. 
     The blind detection circuit of  FIG. 7A , therefore, reliably detects the presence of transmit diversity in less than 250 milliseconds. Moreover, the probability of missing P m  an active diversity antenna is less than 1.7×10 −3 , and the probability of false detection P f  of an absent transmit diversity antenna is less than 6.5×10 −3 . This method of detection is highly advantageous when time permits. No special consideration is required at the base station to accommodate mobile detection. The mobile relies on a ratio of signals from the primary and diversity antennas for detection. Thus, decoding of transmitted signals is unnecessary for this method of blind detection. 
     Turning now to  FIG. 7B , there is a second embodiment of a mobile unit of the present invention configured for Level 3 (L3) message detection of transmit diversity. This L3 message is a QPSK-encoded binary message that is transmitted on the PCCPCH together with other information such as whether the PCCPCH is STTD encoded, neighboring base stations, Secondary Common Control Physical Channel (SCCPCH) offset and base station received power. The mobile unit applies received signals to the delay profile estimator circuit  720  and signal buffer circuit  706  as previously described. The delay profile estimator circuit applies a control signal corresponding to the Doppler rate of the received signal to the WMSA channel estimate circuit  732  via lead  728 . This control signal determines the variable K number of time slots used by the WMSA channel estimate circuit  732  ( FIG. 5 ). The mobile unit initially assumes the received signal is STTD encoded and produces a corresponding diversity control signal on lead  738 . The diversity control signal enables production of channel estimate signals α j   1  and α j   2  at leads  734  and  736 , respectively. Phase correction circuit  710  receives these channel estimate signals together with the data signals on lead  708  and produces a phase-corrected signal at rake combiner circuit  710 . If the received data signal is STTD-encoded, the rake combiner circuit  712  combines multi-path data signals from the respective primary and diversity antennas and applies them to Viterbi decoder circuit  714 . The Viterbi decoder circuit  714  decodes the L3 message to determine if the message contains information on STTD signal encoding of PCCPCH and produces diversity control signal on lead  738 . If the L3 message contains information that the original PCCPCH data was STTD-encoded, operation of the receiver continues as previously described. Thus, the mobile unit with STTD realizes a typical 3 dB gain for a 5 Hz Doppler frequency corresponding to pedestrian indoor-to-outdoor transmission and a typical 0.6 dB gain for a 200 Hz Doppler frequency corresponding to vehicular transmission compared to non-diversity transmission. 
     When the L3 message includes information that the original PCCPCH data was not STTD-encoded, however, the Viterbi decoder circuit  714  changes the logic state of the control signal on lead  738 . This non-diversity control signal on lead  738  disables the diversity channel estimate α j   2  on lead  736 . The non-diversity control signal further disables the phase-corrected output from phase correction circuit  710 , thereby eliminating noise at the rake combiner circuit  712  due to an absent diversity signal. 
     The received L3 message is degraded at the mobile receiver when STTD signal encoding is incorrectly assumed prior to initial decoding. This degradation is due to noise at the rake combiner circuit fingers corresponding to the absent diversity antenna. The degradation due to this noise is shown at the simulated output of  FIG. 10A . The simulation compares normal and STTD signal decoding of single antenna transmission for a Doppler frequency of 5 Hz with weighted multi-slot averaging (WMSA). The received channel energy to noise ratio (E 0 /N 0 ) increases by only 0.2-0.4 dB for a selected bit error rate (BER). A comparable result is evident from the simulation comparing normal and STTD signal decoding of single antenna transmission for a Doppler frequency of 200 Hz with WMSA ( FIG. 10B ). The received channel energy to noise ratio (E 0 /N 0 ) for this vehicular Doppler frequency of 200 Hz increases by 0.6-0.7 dB for a selected bit error rate (BER). A nominal degradation of the received channel energy to noise ratio (E 0 /N 0 ) of 0.2-0.7 dB at the mobile unit will not inhibit correct demodulation of the L3 diversity message. This method of diversity detection is highly advantageous in reducing diversity detection time. The decoding of the information in the L3 message affirmatively indicates the presence or absence of diversity transmission at the output of the Viterbi decoder in less than 30 milliseconds. Only nominal signal degradation occurs by incorrectly decoding a non-diversity L3 message as though it were STTD signal encoded. 
     Although the invention has been described in detail with reference to its preferred embodiment, it is to be understood that this description is by way of example only and is not to be construed in a limiting sense. For example, advantages of the present invention may be achieved by a digital signal processing circuit utilizing a combination of hardware and software operations as will be appreciated by those of ordinary skill in the art having access to the instant specification. Furthermore, the advantages the blind detection method of signal diversity detection of  FIG. 7A  and the L3 message diversity detection of  FIG. 7B  may be combined. For example, the mobile unit may initially use the blind detection method to determine a presence or absence of transmit diversity. The result of this determination may then be used to decode the L3 message from the base station. The decoded L3 message may then be used to confirm the blind detection result. When results differ, however, the process may be repeated. In another embodiment of the present invention, the mobile unit may use either blind detection or L3 message decoding to determine a presence or absence of transmit diversity among neighboring base stations as well as a selected base station. In yet another embodiment of the present invention the mobile unit may receive transmit diversity information together with long code group information about neighboring base stations from the selected base station via L3 message decoding.