Abstract:
A demodulation circuit for recovering frequency redundant quadrature amplitude modulated data comprises an A/D converter generating digitized carrier signal. A mixer receives the digitized carrier and generates a frequency shifted I-signal and a frequency shifted Q-signal. A bank of filters includes a narrow band filter centered at each one of a plurality frequency shifted sub-spectra to generate a digital data signal for each sub-spectrum. A selection circuit determines which one of the sub-spectrum data signals is the strongest and a receiver circuit recovers the transmitted data in response to the strongest sub spectrum data signal.

Description:
TECHNICAL FIELD  
         [0001]    The present invention relates generally to network interfacing, and more particularly, to a method for recovering frequency redundant data in a network communications receiver.  
         BACKGROUND OF THE INVENTION  
         [0002]    The transmission of various types of digital data between computers continues to grow in importance. The predominant method of transmitting such digital data includes coding the digital data into a low frequency baseband data signal and modulating the baseband data signal onto a high frequency carrier signal. The high frequency carrier signal is then transmitted across a network physical transmission medium such as electrical cable, fiber optic, RF, or other medium to a remote computing station.  
           [0003]    At the remote computing station, the high frequency carrier signal must be received and demodulated to recover the original baseband signal. In the absence of any distortion across the network medium, the received signal would be identical in phase, amplitude, and frequency to the transmitted carrier and could be demodulated using known mixing techniques to recover the baseband signal. The baseband signal could then be recovered into digital data using known sampling algorithms.  
           [0004]    One problem with such networks is that the physical medium and network topology tend to distort the high frequency carrier signal. Branch connections and different lengths of such branches cause reflections of the transmitted signal. Such problems are even more apparent in a network which uses home telephone wiring cables as the network physical medium. The typical wiring of the telephone network is designed for the “plain old telephone service” (POTS) signals in the 0.3-3.4 kilohertz frequency and are not designed for transmission of high frequency carrier signals in a frequency range greater than 1 MHz. The high frequency carrier signal is also distorted by transients in wiring characteristics due to on-hook and off-hook switching and noise pulses of the POTS (e.g. ringing). The high frequency carrier is further distorted by spurious noise caused by electrical devices operating in close proximity to the “cable” medium.  
           [0005]    Such distortion of frequency, amplitude, and phase of the high frequency carrier signal degrades network performance and tends to impede the design of higher data rate networks. Known techniques for compensating for such distortion and improving the data rate of a network include complex modulation schemes and frequency diversity.  
           [0006]    Utilizing a complex modulation scheme such as quadrature amplitude modulation (QAM), both the amplitude and phase of the high frequency carrier are modulated to represent in-phase (I) and quadrature (Q) components of a baseband signal. Referring to FIG. 1, a 4-QAM modulation constellation  10  is shown. In operation, each data symbol is represented by an I-value of +1 or −1 and a Q-value of +1 or −1 such that the data symbol can be represented by one of the four states  12 ( a )- 12 ( d ) in constellation  10 . Each constellation point  12 ( a ) - 12 ( d ) represents a unique combination of carrier amplitude and phase. For example, constellation point  12 ( a ) represents a carrier amplitude of  14  and a carrier phase  16 .  
           [0007]    [0007]FIG. 2 illustrates the utilization of frequency diversity by transmitting the same data in three mutually exclusive sub-spectra  18 ( a )- 18 ( c ) of the transmission band  20 . Therefore, if a portion of the band is distorted (e.g. one or more of the sub-spectra  18 ( a )- 18 ( c ), the data may still be recovered at the receiver from a less distorted portion of the sub-spectra  18 ( a )- 18 ( c a). For example, a data signal modulated onto a 7 MHz carrier utilizing 6 MHz of bandwidth may include three mutually exclusive sub-bands  18 ( a )- 18 ( c ) centered at 5 MHz, 7 MHz and 9 MHz.  
           [0008]    One approach to demodulating complex signals is to use filters implemented by digital signal processing (DSP), which provides for a convenient way of varying filter coefficients for each transmission to accommodate carrier distortion as detected in the particular time frame in which the data is being transmitted. Using such approach, an equalizer in the receiver compares the distorted received signal representing a known preamble transmission (prior to the data transmission) to the undistorted waveform of the preamble and determines the appropriate filter coefficients for recovery of the received signal. Such filter coefficients are then used for receiving the data transmission.  
           [0009]    In accordance with DSP technology, the high frequency carrier is typically sampled with an A/D converter at a rate that is at least 4 times that of the carrier frequency. Assuming a carrier frequency on the order of 7 MHz, the sampling rate will be on the order of 30 MHz. A problem associated with processing digital samples at such rates to demodulate a complex modulated carrier, and to process mutually exclusive sub-bands of a frequency diverse system, is that very large and costly DSPs would be required.  
           [0010]    More specifically, a typical receiver used for demodulating a complex modulated carrier in a noisy environment includes an equalizer and a slicer digitally implemented. The equalizer includes various filters with adaptive filter coefficients for reshaping the noise-distorted signal. The slicer maps the equalized signal to a sequence of constellation coordinates to recover the transmitted data. The slicer also feeds back error data (difference between an equalized coordinate and a defined coordinate) to the equalizer for updating the filter coefficients.  
           [0011]    A problem with using such known systems in a frequency diverse environment is that the equalizer may include a 10-15 TAP complex FIR filter to equalize the signal and the error feed back loop in the equalizer may include additional filters of comparable complexity. The multipliers associated with these filters require significant hardware size and complexity. Therefore, simply replicating the equalizer and slicer for each of the sub bands in a frequency diverse environment would result in large and costly DSP circuitry.  
           [0012]    Therefore, based on the industry recognized goals for size and cost reduction, what is needed is a receiver system and method for recovering data from a frequency diverse carrier that does not suffer the disadvantages of known systems.  
         SUMMARY OF THE INVENTION  
         [0013]    A first aspect of the present invention is to provide a demodulation circuit comprising: a) an A/D converter generating a series of samples representing a frequency diverse modulated carrier including redundant data transmitted in a plurality of sub-spectra; b) a mixer receiving the series of samples and generating frequency shifted I-signal and a frequency shifted Q-signal representing redundant data in a plurality of frequency shifted sub-spectra; c) a filter bank including a plurality of narrow band filters, each centered at one of a plurality of the frequency shifted sub-spectra, each receiving the frequency shifted I-signal and the frequency shifted Q-signal, and each generating a sub-spectrum data signal comprising a sub spectrum I-signal and a sub-spectrum Q-signal; d) a selection circuit determining which one of the sub-spectrum data signals is the strongest sub-spectrum data signal; and e) a receiver circuit generating digital data in response to the strongest sub spectrum data signal.  
           [0014]    In the preferred embodiment, the selection circuit operates by comparing a strength value of each sub-spectrum data signal to a strength value of at least one other of the sub-spectrum data signals and selecting the sub spectrum data signal with the greatest strength value to pass to the receiver. The strength value may represent the sum of the strength of a plurality of sample values in the sub-spectrum data signal. The sample values from the first sub-spectrum and the second sub-spectrum may be within a defined time window and the strength value may be equal to the sum of the square of each of an I sample value and a Q sample value occurring within the defined time window.  
           [0015]    Further, the selection circuit may select a plurality of defined time windows within a transmission frame and select sub-spectrum with the largest strength value for each of the defined time windows to pass to the receiver to provide the receiver circuitry with the highness strength signal throughout the transmission frame. Further yet, the receiver may include an equalizer and a slicer for recovering quadrature amplitude modulated data from the sub-spectrum data signal. There may be three 2 Mhz sub-spectra centered at 5 Mhz, 7 Mhz, and 9 Mhz respectively.  
           [0016]    A second aspect of the present invention is to provide a method of recovering data from a frequency diverse modulated carrier comprising: a) generating a series of digital samples representing redundant data transmitted in a plurality of sub-spectra of a frequency diverse modulated carrier; b) mixing the series of digital samples with a digital sine waveform and a digital cosine waveform to generate a frequency shifted I-signal and a frequency shifted Q-signal representing redundant data in a plurality of frequency shifted sub-spectra; c) filtering the frequency shifted I-signal and the frequency shifted Q-signal with a filter bank including a plurality of narrow band filters, each centered at one of a plurality of the frequency shifted sub-spectra to generate a plurality of sub-spectrum data signals, comprising a sub spectrum I-signal and a sub-spectrum Q-signal, each corresponding to one of the frequency shifted sub-spectra; d) determining which one of the sub-spectrum data signals is the strongest sub-spectrum data signal; and e) recovering digital data in response to the strongest sub spectrum data signal.  
           [0017]    The step of determining which one of the sub-spectrum data signals is the strongest sub-spectrum data signal may include comparing a strength value of each sub-spectrum data signal to a strength value of at least one other of the sub-spectrum data signals and selecting the sub spectrum data signal with the greatest strength value. The strength value may represent the sum of the strength of a plurality of sample values in the sub-spectrum data signal. The sample values from the first sub-spectrum and the second sub-spectrum may be within a defined time window and the strength value may be equal to the sum of the square of each of an I sample value and a Q sample value occurring within the defined time window.  
           [0018]    The method may further include determining which one of the sub-spectrum data signals is the strongest sub-spectrum data signal for a plurality of defined time windows within a transmission frame and selecting the sub-spectrum with the largest strength value for each of the defined time windows to provide the receiver circuitry with the highness strength signal throughout the transmission frame. Further, the method may include equalizing and slicing the sub-spectrum data signal with the highest strength to recover quadrature amplitude modulated data from the sub-spectrum data signal. There may be three 2 Mhz sub-spectra centered at 5 Mhz, 7 Mhz, and 9 Mhz respectively.  
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0019]    [0019]FIG. 1 is a diagram of a 4-QAM encoding/decoding constellation useful for the practice of one embodiment of the present invention;  
         [0020]    [0020]FIG. 2 is a frequency domain diagram of a frequency diverse redundant data signal;  
         [0021]    [0021]FIG. 3 is a block diagram of a local area network in accordance with one embodiment of this invention;  
         [0022]    [0022]FIG. 4 is a block diagram of a receiver circuit in accordance with one embodiment of this invention;  
         [0023]    [0023]FIG. 5 is a block diagram of a strength comparison circuit in accordance with one embodiment of this invention;  
         [0024]    [0024]FIG. 6 is a flow chart showing exemplary operation of the circuit if FIG. 5;  
         [0025]    [0025]FIG. 7 a  is a time domain diagram of a digital signal; and  
         [0026]    [0026]FIG. 7 b  is a time domain diagram of a digital signal. 
     
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0027]    The present invention will now be described in detail with reference to the drawings. In the drawings, like reference numerals are used to refer to like elements throughout.  
         [0028]    [0028]FIG. 3 is a diagram of a local area network  40  implemented in a home environment using twisted pair network media according to an embodiment of this invention. The network  40  includes network stations  42 ( a )- 42 ( c ) that are connected to a telephone line (twisted pair) wiring  48  via RJ- 41  phone jacks  44 ( a )- 44 ( c ) respectively. A telephone  46  is connected to an RJ- 11  phone jack  44 ( d ) and may continue to make telephone calls while network devices  42 ( a )- 42 ( c ) are communicating network data.  
         [0029]    Each network device  42  may be a personal computer, printer, server, or other intelligent consumer device and each includes a transceiver  43 ( a )- 43 ( c ) respectively for communicating with other network devices  42 ( a )- 42 ( c ) via the network media  48 .  
         [0030]    Each of network devices  42 ( a )- 42 ( c ) communicates by transmitting an analog network signal comprising a carrier signal modulated with network data. A portion of network devices  42 ( a )- 42 ( c ) may utilize pulse position modulation (PPM) wherein network data is modulated on the carrier utilizing discrete band limited pulses. Another portion of network devices  42 ( a )- 42 ( c ) may selectively utilize PPM modulation or frequency diverse quadrature amplitude modulation (QAM) wherein network data is modulated on the carrier by varying both the amplitude and phase of the carrier in accordance with a complex encoding constellation in multiple sub bands. In the preferred embodiment, the PPM modulation adheres to the Home Phoneline Network Alliance (HPNA) 1.0 standard, as promulgated by a consortium of network equipment providers including Advance Micro Devices, Inc. of Sunnyvale, Calif., and provides for a 1 Mbit data rate and the frequency diverse QAM modulation adheres to the HPNA 2.0 standard and provides for a 10 Mbit data rate. As such, it should be appreciated that data communication between a networked device  42  which is only capable of PPM modulation and any other device will utilize a PPM modulated carrier. However, data communication between two devices that are capable of both PPM and frequency diverse QAM modulation will utilize the faster data rate of a frequency diverse QAM modulated carrier. Therefore, at the receiving network device, it should be appreciated that a transceiver  43 ( a )- 43 ( c ) capable of both PPM and frequency diverse QAM must be able to receive data utilizing both a PPM and a frequency diverse QAM modulated carrier.  
         [0031]    Referring to FIG. 4, a block diagram of a receiver  45  for recovering data from a frequency diverse QAM carrier is shown. Analog front end  50  is coupled to the network wiring  48  and includes known analog circuits tuned to the modulated carrier frequency for receiving a carrier signal from the transmission medium and amplifying such signal to utilize the dynamic range of A/D converter  52 .  
         [0032]    In the preferred embodiment A/D converter  52  is a 10 -bit A/D converter. The A/D converter  52  is driven by a sampling clock signal  60  with a frequency that is at least four-times that of a nominal frequency of the modulated carrier frequency. In the preferred embodiment, the nominal frequency of the modulated carrier is 7 Mhz and a 32 Mhz sampling frequency is chosen.  
         [0033]    The output of the A/D converter  52  is a digitized carrier signal on line  62  comprising a sequence of 10-bit sample values at the 32 Mhz sample frequency. The digitized carrier signal on line  62  is input to a complex mixer  54  which operates to mix the digital sample values with a digital sine wave  64 (I) in mixer  56 (I) and a digital cosine wave  64 (Q) (of the same frequency as the digital sine wave) in mixer  56 (Q). The outputs on lines  58 (I) and  58 (Q) are thus a frequency shifted inphase (I) signal and a frequency shifted quadrature (Q) signal respectively. The frequency shift is equal to the frequency of the sine wave and the cosine wave. Because a sine wave with a frequency of one fourth of the sampling frequency can be digitally represented by a sequence of values comprising only plus/minus 1 and zero, in the preferred embodiment, the digital sine waveform and cosine waveform have a frequency of 8 Mhz to simplify the digital hardware for performing the mixing operation.  
         [0034]    The frequency shifted I signal and Q signal on lines  58 (I) and  58 (Q) are input to a sub band selector  66 . The sub-band selector  66  includes a bank of narrow band filters  68 ( a )- 68 ( c ), each of which passes a frequency band centered at one of the frequency shifted sub spectra while not passing frequencies outside of the sub-spectra. The output from each narrow band filter  68 ( a )- 68 ( c ) is input to signal strength comparator and selector  70 . Signal strength comparator and selector  70  compares the signal strength in each sub spectra and selects the sub spectra with the most powerful signal for output on lines  72 (I) and  72 (Q) for equalizing and slicing for data recovery.  
         [0035]    Equalizer  78  receives the baseband I signal and the baseband Q signal and utilizing an 11TAP FIR filter with adaptive coefficients shapes the signals to accommodate for distortion in network  48 . An equalized I signal and an equalized Q signal are output on lines  80 (I) and  80 ( q ) for data recovery in slicer  82 .  
         [0036]    Slicer  82  operates to map each combination of an I value from the I equalized I signal and a Q value from the equalized Q signal to a defined constellation coordinate and to look-up the payload encoded data associated with such defined coordinate to recover the transmitted data. A parallel to serial converter  84  sequences the recovered data for output on data line  86 . Slicer  82  also generates a feedback signal on line  88  to equalizer  78  which represents the difference between the received constellation coordinate and the defined constellation coordinate to which it mapped. This difference represents the error between the equalized signal and the originally transmitted signal and, as discussed previously, is used by the equalizer  78  to select and update filter coefficients to more effectively equalize the signals received by the equalizer on lines  72 (I) and  72 (Q).  
         [0037]    Referring to FIG. 5 a more detailed block diagram of compare and select circuit  70  is shown. Circuit  70  receives an I and a Q signal from each of the narrow band filters F 1 , F 2  and F 3 . To determine the strength of the signal in each sub-spectra, the circuit  70  calculates the energy of each sample value which is the square root of I 2 +Q 2  and then sums the square of the energy of each sample value during a time window in each of circuits  96 ( 1 ),  96 ( 2 ) and  96 ( 3 ) corresponding to the respective subbands. It should be appreciated that because the sum of the squares of the square roots of I2+Q2 is being calculated, the actual square root never needs to be calculated and the energy for each sub-spectra during the time window can simply be calculated by summing the squares of each I sample value and each Q sample value.  
         [0038]    For example, referring to FIG. 7 a,  a sequence of sample values  100 ( a )  100 ( s ) is shown within a sample time window  102 . Referring to FIG. 7 b , a second sequence of sample values  104 ( a )- 104 ( s ) is shown within a sample time window  106 . It should be appreciated that the sample values  104 ( a )- 104 ( s ) represent a stronger signal than the sample values  100 ( a )- 100 ( s ) as can be quantified by comparing the sum of the squares of sample values  100 ( a )- 100 ( s ) with the sum of the squares of sample values  104 ( a )- 104 ( s ).  
         [0039]    Referring again to FIG. 5, comparison circuit  98  of compare and select circuit  70  receives the energy value from each circuit  96 ( 1 ),  96 ( 2 ), and  96 ( 3 ) and selects the sub-spectra with the greatest energy value and couples the I and Q signal from that corresponding narrow band filter (e.g. F 1 , F 2 , or F 3 ) to the equalizer  78  (FIG. 4) on lines  72 (I) and  72 (Q).  
         [0040]    Referring to FIG. 6, a flowchart of operation of the compare and select circuit  70  of FIG. 5 is shown. The circuit selects a sample time window in which to compare the signals from each filter representing the different sub-spectra at step  90 . At step  92 , for each sub spectra, the comparator squares each I sample value and Q sample value within the time window and sums the squares. At step  94  the comparator simply compares the sum of the square value from each sub spectra and selects the sub-spectra with the largest sum of the square value to be output to the equalizer  78  on lines  72 (I) and  72 (Q) (FIG. 4).  
         [0041]    The above described systems and methods effectively select the strongest signal from a plurality of sub spectra in a frequency redundant data communication system. As such, the transmitted data can be recovered without redundant hardware in a receiver for separately recovering data from each sub-spectra. Although the invention has been shown and described with respect to certain preferred embodiments, it is obvious that equivalents and modifications will occur to others skilled in the art upon the reading and understanding of the specification. The present invention includes all such equivalents and modifications, and is limited only by the scope of the following claims.