Abstract:
An electronic ballast for driving at least one lamp comprising a rectifying circuit operatively connectable to an AC line; a current drawing circuit connected across said rectifying circuit; and an inverter circuit connected to said rectifying circuit that supplies a lamp current to said at least one lamp; wherein said current drawing circuit draws current from said AC line when the instantaneous voltage of said AC line nears zero to reduce the total harmonic distortion of the input current drawn by said ballast.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS  
       [0001]     This application is a continuation of U.S. patent application Ser. No. 09/887,848, filed Jun. 22, 2001 by Robert S. Newman, Jr., et al. entitled “ELECTRONIC BALLAST” which is assigned to the assignee herein and the entire disclosure of which is hereby incorporated by reference herein. 
     
    
     BACKGROUND OF THE INVENTION  
       [0002]     The present invention relates to electronic ballasts and, more particularly, to electronic dimming ballasts for gas discharge lamps, such as fluorescent lamps.  
         [0003]     Electronic ballasts for fluorescent lamps typically can be analyzed as comprising a “front end” and a “back end”. The front end typically includes a rectifier for changing alternating current (AC) mains line voltage to a direct current (DC) bus voltage and a filter circuit for filtering the DC bus voltage. Electronic ballasts also often use a boost circuit for boosting the magnitude of the DC bus voltage. The filter circuit typically comprises a capacitive low-pass filter.  
         [0004]     The ballast back end typically includes a switching inverter for converting the DC bus voltage to a high-frequency AC voltage, and a resonant tank circuit having a relatively high output impedance for coupling the high-frequency AC voltage to the lamp electrodes. The ballast back end also typically includes a feedback circuit that monitors the lamp current and generates control signals to control the switching of the inverter so as to maintain a desired lamp current magnitude.  
         [0005]     In order to maintain stable lamp operation, typical prior art electronic ballasts filter the DC bus voltage to minimize the amount of bus voltage ripple. This is usually accomplished by providing a bus capacitor having a relatively large capacitance and hence, a relatively large energy storage capacity. By providing a relatively large bus capacitor, the amount of decay from the rectified peak voltage is minimized from one half-cycle to the next half-cycle. Minimizing the amount of ripple on the DC bus also tends to minimize the current crest factor (CCF) of the lamp current. The CCF is defined as the ratio of the magnitude of the peak lamp current to the magnitude of the root-mean-square (RMS) value of the lamp current.  
             CCF   ≡       I   pk       I   RMS               (     Equation   ⁢           ⁢   1     )             
 
         [0006]     However, using a relatively large bus capacitor to minimize ripple on the DC bus voltage comes with its disadvantages. The bus capacitor must be fairly large, and is therefore more expensive and consumes more area on a printed circuit board, or the like, and volume within the ballast. Also, because the bus capacitor is discharging whenever the bus voltage level is above the absolute value of the AC mains voltage, the bus capacitor is recharging during a relatively short duration of each line half-cycle centered about the peak of the line voltage. This results in the typical prior art ballast having a tendency to draw a relatively large amount of current during the time that the bus capacitor is charging, as shown in  FIG. 1 . Consequently, this results in a distorted input current waveform giving rise to unwanted harmonics and undesirable total harmonic distortion (THD) and power factor for the ballast.  
         [0007]     One approach to lowering the THD and improving the ballast power factor has been to employ an active power factor correction (APFC) circuit. This approach has resulted in tradeoffs. Using an APFC circuit results in added ballast complexity, more components, greater cost, lower reliability, and possibly increased power consumption. Moreover, the ballast typically uses a relatively large bus capacitor with its attendant disadvantages as noted above.  
         [0008]     Another approach to lowering THD and improving ballast power factor has been to employ a valley-fill circuit between a rectifier and an inverter. One disadvantage of typical prior art valley-fill circuits is that they can have greater bus ripple, which results in higher lamp CCF, which can in turn shorten lamp life.  
         [0009]     Prior art approaches to providing electronic ballasts having improved power factor are and THD are discussed in T.-F. Wu, Y.-J. Wu, C.-H. Chang and Z.-R. Liu, “Ripple-Free, Single-Stage Electronic Ballasts with Dither-Booster Power Factor Corrector”, IEEE Industry Applications Society Annual Meeting, pp. 2372-77, 1997; Y.-S. Youn, G. Chae, and G.-H. Cho, “A Unity Power Factor Electronic Ballast for Fluorescent Lamp having Improved Valley Fill and Valley Boost Converter”, IEEE PESC97 Record, pp. 53-59, 1997; and G. Chae, Y.-S. Youn, and G.-H. Cho, “High Power Factor Correction Circuit using Valley Charge-Pumping for Low Cost Electronic Ballasts”, IEEE 0-7803-4489-8/98, pp. 2003-8, 1998.  
         [0010]     Prior art patents representative of attempts to provide electronic ballasts having improved power factor and THD include U.S. Pat. No. 5,387,847, “Passive Power Factor Ballast Circuit for the Gas Discharge Lamps”, issued Feb. 7, 1995 to Wood; U.S. Pat. No. 5,399,944, “Ballast Circuit for Driving Gas Discharge”, issued Mar. 21, 1995 to Konopka et al.; U.S. Pat. No. 5,517,086, “Modified Valley Fill High Power Factor Correction Ballast”, issued May 14, 1996 to El-Hamamsy et al.; and U.S. Pat. No. 5,994,847, “Electronic Ballast with Lamp Current Valley-Fill Power Factor Correction”, issued Nov. 30, 1999.  
       SUMMARY OF THE INVENTION  
       [0011]     In accordance with a first feature of the invention, a novel electronic ballast for driving a gas discharge lamp includes a rectifier to convert an AC mains input voltage to a rectified pulsating voltage, a valley-fill circuit including an energy storage element to which energy is delivered through a switched impedance to fill the valleys between successive rectified voltage peaks to produce a substantially DC bus voltage, a DC-to-AC voltage inverter having series-connected switching devices to convert the substantially DC bus voltage to a high-frequency AC voltage signal for driving the gas discharge lamp, a resonant tank for coupling the high-frequency AC voltage signal to the gas discharge lamp, and a control circuit for controlling the switching action of the switching devices to deliver a desired current to the gas discharge lamp.  
         [0012]     In a preferred embodiment of the ballast, the energy storage element of the valley-fill circuit includes a bus capacitor that stores energy during a first predetermined charging portion of each half-cycle of the AC mains voltage, and delivers energy to the gas discharge lamp during a second predetermined discharge portion of each half-cycle of the AC mains voltage. The switched impedance of the valley-fill circuit includes a resistor in series with a controllably conductive device (a field effect transistor), through which the bus capacitor is charged.  
         [0013]     In an alternative embodiment, the energy storage device of the valley-fill circuit includes a bus capacitor, and the switched impedance includes a buck inductor in series with a field effect transistor, connected together in a buck converter configuration. The bus capacitor stores energy during a first predetermined charging portion of each half-cycle of the AC mains voltage, and delivers energy to the gas discharge lamp during a second predetermined discharge portion of each half-cycle of the AC mains voltage. The buck inductor stores energy in response to conduction of the field effect transistor during the charging period of the bus capacitor, and transfers the stored energy to the bus capacitor in response to non-conduction of the field effect transistor during the charging period of the bus capacitor.  
         [0014]     In an alternative embodiment, the buck inductor is provided with a tap connected to the bus voltage through a commutation diode to provide different energy storage and energy transfer times.  
         [0015]     In accordance with a second feature of the invention, a novel electronic ballast for driving a gas discharge lamp includes a rectifier to convert an AC mains input voltage to a rectified pulsating voltage, a valley-fill circuit to fill the valleys between successive rectified voltage peaks to produce a substantially DC bus voltage, a DC-to-AC voltage inverter having series-connected switching devices to convert the substantially DC bus voltage to a high-frequency AC voltage signal for driving the gas discharge lamp, a resonant tank for coupling the high-frequency AC voltage signal to the gas discharge lamp, a control circuit for controlling the switching action of the switching devices to deliver a desired current to the gas discharge lamp, and means for drawing input current near the zero crossing of the AC mains input voltage waveform so that the input current THD is substantially reduced, and the power factor of the ballast is increased.  
         [0016]     In a preferred embodiment of the ballast, the means for drawing current near the zero crossing is a cat ear power supply that supplies the power necessary to operate the control circuit. The cat ear power supply draws current from the AC mains from near the zero crossing of the AC mains voltage at either the leading edge of each half-cycle, or the trailing edge of each half-cycle, or both. The cat ear power supply derives its name from the shape of its input current waveform which “fills in” the current waveform drawn by the ballast from the AC mains around the zero crossings. The cat ear power supply may be provided with circuitry that “cuts in” and “cuts out” the power supply in response to fixed input voltage levels. Alternatively, the cat ear power supply may be provided with circuitry to monitor the current drawn by the ballast back end and cause the power supply to draw input current only when the back end is not drawing current. 
     
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0017]      FIG. 1  is a representation of voltage and current waveforms in a prior art electronic ballast with no APFC or valley-fill circuits with idealized waveforms shown as dashed lines;  
         [0018]      FIG. 2  is a simplified block diagram of the electronic ballast of the invention;  
         [0019]      FIG. 3  is a simplified schematic diagram of a first embodiment of a valley-fill circuit using a buck converter for use in an electronic ballast of the invention;  
         [0020]      FIG. 4  is a simplified representation of bus voltage in the buck converter of  FIG. 3  illustrating the method of operation;  
         [0021]      FIG. 5  is a simplified schematic of the buck converter of  FIG. 3  illustrating a first mode of operation;  
         [0022]      FIG. 6  is a simplified schematic of the buck converter of  FIG. 3  illustrating a second mode of operation;  
         [0023]      FIG. 7  is a simplified representation of various voltage and current waveforms in an electronic ballast including the buck converter of  FIG. 3  at full light output;  
         [0024]      FIG. 8  is a simplified representation of various voltage and current waveforms in an electronic ballast including the buck converter of  FIG. 3  at ten percent light output;  
         [0025]      FIG. 9  is a simplified schematic diagram of a second embodiment of a valley-fill circuit having a buck converter integrated with an inverter in accordance with the present invention;  
         [0026]      FIG. 10  is a simplified schematic diagram of a third embodiment of a valley-fill circuit having an integrated buck converter with a tapped buck inductor in accordance with the invention;  
         [0027]      FIG. 11  is a simplified schematic diagram of another alternative embodiment of a valley-fill circuit having a flyback transformer for recharging the bus capacitor;  
         [0028]      FIG. 12  is a simplified schematic diagram of a fourth embodiment of a valley-fill circuit in accordance with the present invention;  
         [0029]      FIG. 13  is a simplified schematic diagram of a fifth embodiment of a valley-fill circuit in accordance with the present invention;  
         [0030]      FIG. 14  is a simplified schematic diagram of a sixth embodiment of a valley-fill circuit integrated with an inverter in accordance with the present invention;  
         [0031]      FIG. 15  is a simplified schematic diagram of a seventh embodiment of a valley-fill circuit in accordance with the present invention;  
         [0032]      FIG. 16  is a simplified schematic diagram of an eighth embodiment of a valley-fill circuit in accordance with the present invention;  
         [0033]      FIGS. 17 and 18  are simplified schematic diagrams of a ballast constructed in accordance with the invention;  
         [0034]      FIG. 19  is a simplified diagram illustrating the duty cycle of the inverter switches changing over a half-cycle of the line voltage, and the resultant line current drawn by the ballast;  
         [0035]      FIGS. 20 and 21  are simplified schematic diagrams of a second embodiment of an electronic ballast constructed in accordance with the invention;  
         [0036]      FIG. 22  is a simplified partial schematic diagram of the ballast of  FIGS. 20 and 21  showing details of the control, waveshaping, and feedback circuitry;  
         [0037]      FIG. 23  is a simplified schematic diagram of an automatic gain control circuit of the waveshaping circuit of  FIG. 22 ;  
         [0038]      FIG. 24  is a simplified block diagram of a second embodiment of the feedback circuit of  FIG. 20 ;  
         [0039]      FIG. 25  is a simplified block diagram of a third embodiment of the feedback circuit of  FIG. 20 ;  
         [0040]      FIG. 26  is a simplified flowchart illustrating the operation of the feedback circuits of  FIGS. 24 and 25 ;  
         [0041]      FIG. 27  is a simplified schematic of a prior art cat ear power supply;  
         [0042]      FIG. 28  is a simplified waveform diagram illustrating the line current drawn by the cat ear power supply of  FIGS. 20 and 22 ;  
         [0043]      FIG. 29  is a simplified schematic diagram of a first embodiment of a cat ear power supply having fixed cut-in and cut-out points in accordance with the invention;  
         [0044]      FIG. 30  is a simplified schematic diagram of a second embodiment of a cat ear power supply including active monitoring of back end current; and  
         [0045]      FIG. 31  is a simplified waveform diagram illustrating the line current drawn by the electronic ballast of  FIGS. 20 and 21 . 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0046]     The foregoing summary, as well as the following detailed description of the preferred embodiments, is better understood when read in conjunction with the appended drawings. For the purposes of illustrating the invention, there is shown in the drawings an embodiment that is presently preferred, in which like numerals represent similar parts throughout the several views of the drawings, it being understood, however, that the invention is not limited to the specific methods and instrumentalities disclosed.  
         [0000]     Ballast Overview  
         [0047]     Referring first to  FIG. 2 , there is shown a simplified schematic diagram of an electronic ballast  810  constructed in accordance with the invention. The ballast  810  includes a rectifier  820  capable of being connected to an AC power supply such as a typical 60 Hz AC main. The rectifier  820  converts the AC input voltage to a rectified pulsating DC voltage. The rectifier  820  is connected to a novel valley-fill circuit  830 , to be described, through a diode  840 . A high-frequency filter capacitor  850  is connected across the inputs to the valley-fill circuit  830 . The valley-fill circuit  830  selectively charges and discharges an energy-storage device so as to reduce the amount of bus voltage ripple. The outputs of the valley-fill circuit  830  are in turn connected to the inputs to an inverter  860 . The inverter  860  converts the rectified DC voltage to a high-frequency AC voltage. The outputs of the inverter  860  are connected to an output circuit  870 , which typically includes a resonant tank, and may also include a coupling transformer. The output circuit filters the inverter  860  output to supply essentially sinusoidal voltage, as well as provide voltage gain and increased output impedance. The output circuit  870  is capable of being connected to drive a load  880  such as a gas discharge lamp; for example, a fluorescent lamp. An output current sense circuit  890  coupled to the load  880  provides load current feedback to a control circuit  882 . The control circuit  882  generates control signals to control the operation of the valley-fill circuit  830  and the inverter  860  so as to provide a desired load current to the load  880 . A power supply  884  is connected across the outputs of the rectifier  820  to provide the necessary power for proper operation of the control circuit  882 .  
         [0000]     The Valley-Fill Circuit  
         [0048]     Turning now to  FIG. 3 , there is shown a schematic diagram of a first embodiment  910  of the valley-fill circuit  830  of  FIG. 2  in the form of a buck converter. Connected across first and second input terminals  912  and  914  is an energy-storage device  916 , in the form of a bus capacitor, in series with a first diode  918 . The function of the buck converter  910  is to provide a controlled charging current for the bus capacitor  916 . Connected to the junction of the bus capacitor  916  and the cathode of the first diode  918  is a buck inductor  920  which is connected in series with a second (optional) diode  922  and a controllably conductive switch  924  to a circuit common. Switch  924  is shown as a metal-oxide-silicon field-effect transistor (MOSFET), but may be a bipolar junction transistor (BJT), insulated gate bipolar transistor (IGBT), or other controllably conductive switching device. The buck converter  910  also includes a third commutating diode  926 , which may also be a suitably controlled synchronous rectifier or FET, connected between the junction of the buck inductor  920  and the second diode  922 , and the end of the bus capacitor  916  connected to the input  912 . A first output terminal  928  is connected to input terminal  912 , bus capacitor  916 , and the cathode of commutating diode  926 . A second output terminal  930  is connected to the second input terminal  914 , circuit common, the anode of diode  918 , and the switch  924 .  
         [0049]     The operation of the buck converter  910  will be described in connection with  FIGS. 4, 5 , and  6 . The buck converter  910  operates under two different conditions. In condition I (interval I in  FIG. 4 ), the instantaneous rectified line voltage  1010  applied to the buck converter  910  is equal to or less than the voltage  1012  across the bus capacitor  916 , and the bus capacitor  916  is discharging its stored energy into the inverter. In this condition, the diode  840  ( FIG. 2 ) is reverse biased, and the diode  918  is forward biased into conduction. This establishes a discharge path for the bus capacitor  916  from circuit common  930 , through the diode  918  and the bus capacitor  916 , to the buck converter output  928 . Switch  924  alternately opens and closes at a frequency substantially greater than the frequency of the rectified line voltage, typically about 30 kHz or greater. When the switch  924  is conducting, any residual energy remaining in the buck inductor  920  from the previous charging cycle is discharged through the diode  922  and the switch  924  to circuit common. Thereafter, the diodes  922  and  926  are reverse biased so that no further current flows through the buck inductor  920 .  
         [0050]     In condition II (interval II in  FIG. 4 ), the instantaneous rectified line voltage is greater than the voltage across the bus capacitor  916 , and the bus capacitor  916  is increasing its stored energy. During interval II, the operation of the buck converter depends on the conduction state of the switch  924 .  
         [0051]     When the switch  924  is conducting, the buck converter  910  reduces to the simplified form shown in  FIG. 5  and the voltage across the buck inductor  920  is equal to the instantaneous rectified line voltage minus the voltage across the bus capacitor  916 . Thus, the bus capacitor  916  is charged by a current flowing from the input  912 , through the bus capacitor  916 , the buck inductor  920 , and the switch  924 , to circuit common. When the switch  924  is not conducting (as depicted in  FIG. 6 ), then the current  1210  flowing through the buck inductor  920  commutates through the diode  926  and flows into the bus capacitor  916 , thus transferring some or all of the energy stored in the inductor  920  to the bus capacitor  916 . Note that in condition II, the bus capacitor  916  is charging both when switch  924  is closed, and when switch  924  is open.  
         [0052]     The result of the operation of the buck converter  910  is that the bus capacitor  916  charges over a longer period of time  1310 , as shown in  FIG. 7  wherein the ballast is operating at full light output. The charging of the bus capacitor  916  preferably takes place over more than 90 degrees of each line half cycle. More preferably, the bus capacitor  916  charges for more than 100 degrees of each half cycle. However, as the charging time increases, so too does the bus voltage ripple. Another advantage of the buck converter valley-fill circuit is that the inrush current to the bus capacitor  916  at the beginning of each charge cycle is limited by the buck inductor  920 . This can also be seen in  FIG. 7  in that the peak line current  1312  is greatly reduced as compared to the peak line current  1314  of typical prior art ballasts having no APFC or valley-fill circuitry. Inrush limiting is even more significant at initial ballast turn-on.  
         [0053]     Another advantage of the buck converter  910  is that it provides over-voltage protection for the bus capacitor  916 . That is, in an unloaded condition, such as when no lamp is present, the bus capacitor  916  will charge to no more than the peak rectified line voltage. This is in contrast to traditional boost and buck-boost converters wherein extra circuitry must be added to prevent charging of the bus capacitor to potentially catastrophically high voltages in an unloaded condition. As shown in  FIG. 8 , as the lamp is dimmed down to about 10 percent light output, the charging time of the bus capacitor  916  decreases while the ballast input current approaches a smoother, more sinusoidal shape. Simultaneously, the bus ripple is also reduced, leading to lower lamp CCF.  
         [0054]     Turning now to  FIG. 9 , there is shown a second embodiment  1410  of the buck converter. In this embodiment, the buck converter  1410  and the inverter  860  share the controllably conductive switch  924 . The second embodiment  1410  of the buck converter operates essentially in the same way as the first embodiment  910 . Turning now to  FIG. 10 , there is shown a third embodiment  1510  of the buck converter in which the buck inductor  920  is replaced with a tapped inductor  1520  and the anode of the commutation diode  926  is coupled to the inductor  1520  tap, rather than the junction of the inductor  1520  and the diode  922 . The inductor tap provides the capability to vary the discharge time of the inductor  1520 . The continuous mode operation of the buck converter may be reduced, or eliminated altogether. However, this extra flexibility comes with the tradeoff of extra voltage stress on the switch  924 . When the inductor  1520  is transferring energy to the bus capacitor  916 , the tapped inductor  1520  acts so that the voltage applied across the switch  924  is equal to the voltage across the capacitor  916  multiplied by the turns ratio of the tapped inductor  1520 . A snubber circuit including a snubber diode  1552 , in series with the parallel combination of a snubber resistor  1554  and a snubber capacitor  1556 , is coupled between the junction of the buck inductor  1520  and the diode  922 , and circuit common, to dissipate uncoupled residual energy in the tapped inductor.  
         [0055]     In the tapped buck inductor embodiment of  FIG. 10 , capacitor  916  is a parallel combination of two 47 microfarad, 250 volt capacitors, diodes  918  and  926  are MUR160 diodes, diodes  922  and  1552  are 1000 volt, 1 ampere diodes, resistor  1554  is a series combination of two 91 kilohm, 1 watt resistors, capacitor  1556  is a 0.0047 microfarad, 630 volt capacitor, and switch  924  is a 500 volt IRF1634G MOSFET. The tapped inductor  1520  has a total number of about 180 turns from the cathode of diode  918  to the anode of diode  922 , corresponding to an inductance of about 1.427 milliHenries, with the number of turns from the cathode of diode  918  to the tap being about 75, corresponding to an inductance of about 244 microHenries, and the number of turns from the tap to the anode of diode  922  being about 105, corresponding to an inductance of about 492 microHenries.  
         [0056]     In each embodiment of the valley-fill circuit  830  ( FIG. 2 ), charging current increases with longer conduction times of the controllably conductive switch. This means that at low end, when the lamp is being dimmed, and the switch is conducting longer, then the accumulated charge is greater, which tends to raise the bus voltage. It is advantageous to have higher bus voltage at low end since the lamp voltage increases at low end and a higher bus voltage allows the lamp to be driven through a higher impedance. Higher output impedance improves lamp stability, as is discussed in U.S. Pat. No. 5,041,763, issued Aug. 20, 1991 to Sullivan et al., and assigned to Lutron Electronics Co., Inc.  
         [0057]     The charging current also increases as the voltage differential between the rectified line voltage and the voltage across the bus capacitor increases. This results in the charging current in the buck converter being highest in the middle of the line half cycle, and lower towards the tails of the line half cycle, which in turn results in improved power factor and THD for the ballast.  
         [0058]     Turning now to  FIG. 11 , there is shown another embodiment  1570  of a valley-fill circuit. In this embodiment, the valley-fill circuit  1570  includes, in addition to the bus capacitor  916 , the diode  922 , and the switch  924 , a diode  1572  connected between the capacitor  916  and the terminal  912 , and a diode  1574  and a “flyback” transformer  1576 . The “primary” winding of the transformer  1576  is connected between the anode of the diode  922  and the terminal  928  of the valley-fill circuit  1570 . The “secondary” winding of the transformer  1576  is connected between circuit common and to the anode of the diode  1574 , the cathode of which is in turn connected to the junction of the capacitor  916  and the anode of the diode  1572 .  
         [0059]     When the rectified line voltage exceeds the voltage across the bus capacitor  916 , then the voltage developed across the “secondary” winding of the flyback transformer  1576  recharges the bus capacitor  916  through the diode  1574  when the “secondary” winding voltage exceeds the bus capacitor voltage. When the rectified line voltage falls below the voltage across the bus capacitor, then the bus capacitor  916  alternate discharges when the “secondary” winding voltage is lower than the bus capacitor voltage, and recharges when the “secondary” winding voltage exceeds the bus capacitor voltage.  
         [0060]     Turning now to  FIG. 12 , there is shown a fourth embodiment  1610  of a valley-fill circuit. In this embodiment, the valley-fill circuit  1610  includes a first energy storage capacitor  1632  connected in series with a first diode  1634  across first and second input terminals  912 ,  914  to the circuit  1610 . A second energy storage capacitor  1616  is connected in series with a second diode  1636 , the cathode of which is tied to the input  912 . A third diode  1638  is connected between the junction of capacitor  1632  and diode  1634 , and the junction of capacitor  1616  and diode  1636 . The other end of energy storage capacitor  1616  is connected to the second input terminal  914  by means of a fourth diode  1618  in parallel with a resistor  1620 . When the rectified voltage across the input terminals  912 ,  914  exceeds the sum of the voltages across the capacitors  1632  and  1616  by the forward voltage drop across the diode  1638 , then the diodes  1634 ,  1636 , and  1618  are reverse biased, the diode  1638  is forward biased, and the energy storage capacitors  1632 ,  1616  charge through the series path of capacitor  1632 , diode  1638 , capacitor  1616 , and resistor  1620 . Resistor  1620  limits the inrush charging current into the energy storage capacitors  1632 ,  1616  to reduce current spikes in the current drawn by the ballast from the line, thereby improving power factor and THD. The capacitors  1632 ,  1616  typically each have the same value and charge to approximately one-half the peak input voltage. When the rectified voltage across the input terminals  912 ,  914  falls below the sum of the voltage across capacitors  1632 ,  1616 , then diode  1638  is reverse biased. Once the voltage across the input terminals  912 ,  914  falls below the voltage across the capacitor  1632  by more than the turn-on voltage of the diode  1634 , then capacitor  1632  discharges through diode  1634  and output terminals  928  and  930 . Once the voltage across the input terminals  912 ,  914  falls below the voltage across the capacitor  1616  by more than the turn-on voltage of the diode  1636 , then capacitor  1616  discharges through the diode  1636 , the resistor  1620 , and the output terminals  928 ,  930 . When the voltage drop across the resistor  1620  exceeds the turn-on voltage of the diode  1618 , then capacitor  1616  discharges through diodes  1636 ,  1618  and the output terminals  928 ,  930 .  
         [0061]     In summary, the capacitors  1632 ,  1616  charge in series, and discharge in parallel, delivering their stored energy to the lamp. The amount of ripple in the bus voltage is thereby reduced, which in turn leads to improved current crest factor in the ballast.  
         [0062]     The valley-fill circuit  1610  of  FIG. 12  differs significantly from the valley-fill circuit of Wood in U.S. Pat. No. 5,387,847. Most notably, Wood, in  FIG. 2  of his patent, shows a resistor in series with a diode connected between two capacitors. In contrast, the valley-fill circuit of  FIG. 12  provides the resistor  1620  in parallel with the diode  1618 , the pair connected between the capacitor  1616  and circuit common. This novel arrangement provides the desired degree of power factor improvement, but does so in a manner that more readily lends itself to additional improvements in power factor.  
         [0063]     To further improve the power factor of the ballast, the valley-fill circuit  1610  of  FIG. 12  may be modified by adding a controllably conductive switch  924  in series with the resistor  1620  as shown in  FIG. 13 . The switch  924  will typically be operated at a high frequency, that is, many times greater than the fundamental line frequency. The power factor of the ballast may be improved by controlling the switching action of switch  924  so that the conduction time of the switch  924  is reduced near the “tails” or zero crossings of the line voltage, and increased near the center or peak of each line half cycle. This results in a ballast input current waveform that more nearly matches the line voltage waveform.  
         [0064]     The valley-fill circuit  1710  of  FIG. 13  may be integrated with the ballast inverter as shown in  FIG. 14  wherein the controllable switch  924  is shared by the valley-fill circuit  1810  and the inverter.  
         [0065]     Resistive losses in the valley-fill circuit  1710  of  FIG. 13  may be reduced as shown in  FIG. 15  by replacing resistor  1620  with an inductor  1920  in series with switch  924 . In an alternative arrangement, the combination of the inductor  1920  and the switch  924  may be replaced by a single large inductor. However, the high-frequency switching action of switch  924  permits a relatively small, and inexpensive, inductor  1920  to be used.  
         [0066]     As an alternative to switch  924 , a secondary winding  2024  from a high-frequency transformer may be substituted as shown in  FIG. 16 . High-frequency transformers are typically present in ballasts. By the addition of the proper number of turns of a secondary winding (preferably to an already existing transformer), a voltage of alternating polarity may be introduced in series with inductor  1920 , alternately opposing and aiding the flow of current through the inductor  1920 . The winding  2024  thereby effectively functions as a switch. The high-frequency “switching” action of the winding  2024  causes the inductor  1920  to present a higher effective impedance to the current flowing therethrough, thereby allowing a relatively small-valued inductor to be used.  
         [0000]     The Inverter  
         [0067]     As can be seen in  FIGS. 17 and 18 , connected to the output of the bus capacitor  916  and the buck converter  1510  is a high-frequency DC-to-AC inverter  2110  for providing a high-frequency AC signal for driving a gas discharge lamp. The inverter  2110  includes series-connected first and second switching devices  2112  and  924 . The bus voltage is the greater of either the rectified line voltage, or the voltage across the bus capacitor  916 . When the input line voltage is greater than the voltage on the bus capacitor  916 , then the inverter  2110  draws current directly from the AC line. When the rectified line voltage is less than the bus capacitor  916  voltage, then the inverter draws current from the capacitor  916 .  
         [0068]     The operation of the inverter  2110  will now be described in connection with  FIG. 19 . The inverter  2110  uses a fixed frequency, D(1-D) complementary duty cycle switching mode of operation. This means that one, and only one, of the switching devices  2112 ,  924  is conducting at all times. In this discussion, the duty cycle D refers to the conduction time of the first switch  2112 , and the complementary duty cycle 1-D refers to the conduction time of the second switch  924 . When switch  2112  (referred to in  FIG. 19  as SW 1 ) is conducting, then the output of the inverter  2110  is pulled upwardly toward the bus voltage. When the switching device  924  (referred to as SW 2  in  FIG. 19 ) is conducting, then the output of the inverter  2110  is pulled downwardly toward circuit common. Maximum output is achieved when the duty cycle D is equal to 0.5, that is, when the conduction times of the two switching devices  2112 ,  924  are equal. This occurs near the tails of each line voltage half-cycle, that is, near the line voltage zero crossings. The conduction times of the switching devices  2112 ,  924  are controlled by a control circuit  882  in response to the current flowing through the gas discharge lamps  2210 ,  2212 . The operation of the control circuit is described in detail below. The control circuit  882  drives the inverter switches  2112 ,  924  in a manner so that more line current is drawn by the ballast near the peak of each line half-cycle, and less current is drawn near the zero crossings. A more nearly sinusoidal input current waveform is achieved, as shown in  FIG. 19 . This results in improved power factor and input line current THD.  
         [0000]     The Resonant Tank  
         [0069]     Referring again to  FIGS. 17, 18 , the output of the inverter  2110  is connected to a resonant tank circuit  2220  comprising an inductor  2222  and a capacitor  2224 . The resonant tank  2220  filters the inverter  2110  output voltage to supply essentially sinusoidal voltage to the lamps  2210 ,  2212 . In addition, the resonant tank  2220  provides voltage gain and increased output impedance. The output of the resonant tank  2220  is coupled to the electrodes of the gas discharge lamps  2210 ,  2212  by means of a transformer  2230 . A DC blocking capacitor  2232  prevents DC current from flowing through the primary windings of the transformer  2230 .  
         [0000]     The Current Sense Circuit  
         [0070]     Referring to  FIG. 18 , the ballast also includes a current sense circuit  2240  comprising first and second diodes  2242  and  2244 , and resistor  2246 , coupled in series with the lamps  2210 ,  2212 . The current sense circuit  2240  generates a half-wave rectified voltage that is proportional to lamp current and represents a measure of actual light output. The half-wave rectified voltage is supplied as an input to the control circuit  882  of  FIG. 17 . In an alternative embodiment, the current sensing may be performed in a well-known manner by using a current transformer, or alternatively, a full-wave rectification diodes. For non-dimming ballasts, and dimming ballasts where only modest performance is required, the current sense circuit may be omitted.  
         [0000]     The Control Circuit  
         [0071]     The control circuit  882  will be described with reference to  FIGS. 20, 21 , and  22 . A first embodiment of the control circuit  882  generates output drive signals to control the conduction of the switching devices  2112  and  924 . The control circuit  882  receives as an input the half-wave rectified voltage from the current sense circuit  2240  and generates a DC voltage that represents actual light output from the lamps. This DC voltage, representative of light output, is compared to a reference voltage, indicative of a desired light level, to adjust the duty cycles of the switching devices  2112 ,  924  so as to minimize the difference between the light output voltage and the reference voltage. In a dimming electronic ballast, the reference voltage may be provided by an external input such as a 0-to-10 Volt control signal. Alternatively, the reference voltage may be generated by detecting a phase angle control signal applied to the ballast by means of the AC line voltage. In yet another embodiment, the reference voltage may be generated from a phase angle control signal applied to the ballast via an additional input to the ballast, such as is depicted in  FIGS. 20, 22  by the “Dimmed Hot” input.  
         [0072]     The control circuit includes a feedback circuit  2440  connected to receive inputs from the current sense circuit  2240  and a control input circuit  2460 , and supplies drive signals to the gates of the switches  2112  and  924 . The control circuit may optionally include a waveshaping circuit  2480  to provide an additional input to the feedback circuit  2440 , as will be described in detail below.  
         [0073]     As can be seen in  FIG. 22 , the feedback circuit  2440  includes a differential amplifier  2442  connected to receive at its inverting terminal  2444  the input signal representative of lamp light output from the current sense circuit  2240 , and to receive at its non-inverting terminal  2446  the desired light level reference signal. The differential amplifier  2442  produces an error signal representative of the difference between the actual light output and the desired light output. The error signal is in turn provided to a pulse-wave modulation (PWM) circuit  2448  that generates the drive signals that are applied to the gates of the inverter switches  2112  and  924 . The PWM circuit  2448  is well-known in the art and hence will not be described in detail here.  
         [0074]     The control circuit may be designed to have either a “fast” response, or a “slow” response. That is, if the control circuit has a fast response, then it will react quickly to changes in lamp drive current to alter the duty cycles of the inverter switches  2112 ,  924  so as to adjust the output voltage of the inverter  2110  to bring the lamp drive current back to the desired value. While a control circuit having a fast response improves the crest factor of the lamp drive current, this improvement comes at the expense of increased THD for the ballast input current.  
         [0075]     If the control circuit has a slow response, then it will react slowly to changes in lamp drive current. A control circuit having a slow response improves the ballast input current THD, but at the expense of increased lamp drive current crest factor. Striking the appropriate balance between having a control circuit with a fast response or a slow response can be difficult.  
         [0076]     The optional waveshaping circuit  2480  provides an additional means for controlling the rate and manner of response of the control circuit. The waveshaping circuit  2480  provides an AC reference voltage signal that is summed with the essentially DC reference voltage signal from the control input circuit  2460 . While the shape of the AC reference voltage signal can be made to take on a variety of waveshapes depending upon the desires of the circuit designer and the complexity of the waveshaping circuit, a particularly effective, yet simple, circuit can be designed that takes advantage of the waveforms already present in the ballast.  
         [0077]     The waveshaping circuit  2480 , shown in detail in  FIG. 22 , includes a voltage divider including a resistor  2482  connected in series with an automatic gain control (AGC) circuit  2690  that provides a scaled version of the bus voltage from the buck converter  1510 . Details of the AGC  2690  are shown in  FIG. 23  and discussed below. If it is not necessary to adjust the gain of the waveshaping circuit  2480 , such as in non-dimming ballasts, then the AGC  2690  may optionally be replaced by a passive impedance element, such as a resistor.  
         [0078]     The scaled voltage signal from the divider is clipped by means of a diode  2486  having its anode connected to the output of the divider and its cathode connected to a DC reference voltage VREF. The clipped signal is then passed through a DC blocking capacitor  2488  to be summed with the DC reference voltage from the control input circuit  2460 .  
         [0079]     The control circuit also includes a low end clamp  2680  connected between the common tie point for the control input, waveshaping, and feedback circuitry, and circuit common. The low end clamp  2680  prevents the reference voltage from going so low that the current through the lamp cannot be sustained. This prevents the lamp from having to be re-struck each half-cycle.  
         [0080]     The addition of the AC reference signal has the effect of reducing the combined reference voltage when the bus voltage is lower, such as near the zero crossings of the input line voltage, and increasing the combined reference voltage as the bus voltage is increasing, such as when the input line voltage is approaching a peak. The desired lamp current supplied to the lamp by the inverter  2110  will likewise be less when the bus voltage is lower, and increasing when the bus voltage is increasing. Thus, the addition of the AC reference signal, which tracks or follows the bus voltage, has the effect of shaping the current drawn by the lamp to the bus voltage, thereby improving the power factor and THD of the line current drawn by the ballast. However, this improvement in power factor and THD comes at the expense of greater lamp CCF.  
         [0081]     An additional feature of the waveshaping circuit  2480  is the diode  2486  for clipping the peaks of the AC reference signal. During the time when the AC reference voltage signal is clipped, the combined reference voltage remains constant while the bus voltage is peaking. The overall response of the control circuit is designed to be “fast”, so the control circuit responds quickly during the peaks in the bus voltage so as to reduce the duty cycle of switch  2112  so as to deliver a constant voltage, and thus, a constant current, to the lamp. The net effect is to “shave off” the peaks of the lamp drive current envelope by reducing the duty cycle of the switch  2112  so as to reduce peak lamp drive current, and thus reduce lamp CCF. At the same time, the duty cycle of switch  924  increases, thereby increasing the charging time, and hence, the charging current drawn by the bus capacitor  916 , so as to keep the line current drawn by the ballast more nearly sinusoidal so as to reduce ballast input current THD. Electronic dimming ballasts constructed with the waveshaping circuit as described have achieved stable operation with ballast input current THD below 20 percent and lamp drive current crest factor below 1.7.  
         [0082]     The AGC circuit  2690 , shown in  FIG. 23 , reduces the gain of the waveshaping circuit  2480  when it is desired to dim the lamp. The AGC circuit  2690  includes first and second transistors  2691  and  2692 , resistors  2693 ,  2694 , and  2695 , and diode  2696 . The conduction of the first transistor  2691  is controlled by the output of the control input  2460  ( FIG. 22 ). When the input voltage goes low, indicating a dimming condition, conduction of the first transistor  2691  decreases, raising the voltage at the base of the second transistor  2692 , thereby causing the second transistor  2692  to become more conductive, effectively lowering the impedance of the AGC circuit  2690  presented to the waveshaping circuit  2480 , resulting in lower gain for the sampled waveform. Raising the input voltage, such as when it is desired to increase the light level, causes the first transistor  2691  to become more conductive, lowering the voltage at the base of the second transistor  2692 , thereby causing the second transistor  2692  to become less conductive, thereby causing the AGC circuit  2690  to present an effectively greater impedance to the waveshaping circuit  2480 , thereby increasing the gain of the sampled waveform. A second embodiment of the feedback circuit  2440  of  FIG. 20  is shown in  FIG. 24  and includes a microcontroller  26102  coupled to receive as inputs representative of the desired light level and the lamp current, and produce as outputs signals for driving the gates of the inverter switches. One such microcontroller suitable for use is manufactured by Motorola Corporation under the model number MC68HC08. For simplicity, analog-to-digital and digital-to-analog circuits necessary for interfacing the microcontroller  26102  with the analog circuitry of the ballast are considered to be within the ordinary skill of the art and are not shown here.  
         [0083]     A third embodiment of the feedback circuit  2440  of  FIG. 20  is shown in  FIG. 25  and includes in addition to the microcontroller  26102  a gate driver circuit  26104  that receives a single gate drive signal from the microcontroller  26102  and produces signals capable of controlling the operation of the inverter switches. One such gate driver circuit suitable for use is manufactured by International Rectifier under the part number IR2111. Of course, other suitable microcontrollers (such as a PIC16C54A from Power Integrations) and gate drivers may be substituted for the specific embodiments mentioned here. In addition, an application specific integrated circuit (ASIC) (not shown) may be substituted to provide the same functionality as the microcontrollers disclosed herein.  
         [0084]     A high-level flowchart illustrating the operation of the feedback control circuit embodiment of  FIGS. 24 and 25 , shown in  FIG. 26 , includes the steps of measuring the lamp current IL (step  26110 ), and measuring the dimming signal VDIM representative of the desired light level (step  26120 ). The measured lamp current IL is compared to the measured dimming signal VDIM (step  26130 ) and, if IL is less than VDIM, then the duty cycle D of the inverter is increased by increasing the pulse width of the gate drive signal (step  26140 ). If IL is greater than VDIM, as determined in step  26150 , then the duty cycle D is decreased by decreasing the pulse width of the gate drive signal (step  26160 ). If IL is equal to VDIM, then the duty cycle remains unchanged and the process repeats.  
         [0000]     The Cat Ear Power Supply  
         [0085]     Cat ear power supplies have been used for years to provide power for control circuits in two-wire, triac-based dimmers for incandescent lamps and fan motors. A typical prior art cat ear power supply is shown in  FIG. 27 . In such applications, the dimmer is located between the AC mains and the load, receiving as input sinusoidal voltage and providing as output “truncated” form of the sinusoidal input voltage in which the leading edge of the input voltage waveform is blocked by the non-conducting triac, and only the trailing portion of the input voltage waveform is passed on to the load by the triac which is turned on at a predetermined time and conducts until the next zero crossing of the input voltage waveform. By varying the phase conduction angle of the triac, the amount of power delivered to the load may be controlled. The prior art cat ear power supply draws power from the AC mains during a portion of the input voltage waveform when the triac is not conducting. In other words, the prior art cat ear power supply draws current from the line during the time that the load is “off line”. However, until now, cat ear power supplies are not believed to have been used for the purpose of deliberately shaping the current drawn from the line by an electronic ballast (or other electronic devices, such as switch-mode power supplies or line-to-DC converters) so as to improve the power factor and THD of the ballast (or other electronic device) by rendering the current drawn by the ballast more nearly sinusoidal.  
         [0086]     The novel ballast of the invention includes a cat ear power supply  884  ( FIG. 20 ) connected across the outputs of the rectifier  820 . The cat ear power supply may be generally defined as a power supply that is designed to draw current from the line during predetermined portions of the line cycle. The cat ear power supply may thus be used for shaping the ballast input current waveform so as to improve ballast power factor and ballast input current THD. Indeed, the cat ear power supply may be used for shaping the input current waveform of a variety of electronic devices, such as the previously mentioned switch-mode power supplies and line-to-DC converters, for the purpose shaping the input current waveform, improving power factor, and reducing input current THD.  
         [0087]     The cat ear supply  884  ( FIG. 2 ) draws current from the rectifier  820  only at the “tails” of the input line cycle, that is, the regions of the input line cycle adjacent to the line voltage zero crossings, as shown in  FIG. 28 . The cat ear power supply  884  “fills in” the tails of the input line current drawn from the AC mains by the ballast when the back end of the ballast is not drawing current from the line. By filling in the tails, the line current drawn by the ballast is made more nearly sinusoidal, thereby reducing input line current THD, and improving power factor.  
         [0088]     In a first embodiment  2810  of the cat ear power supply  884 , shown in  FIG. 29 , the cat ear power supply  2810  is designed with fixed voltage cut-in and cut-out points. That is, the first embodiment  2810  of the cat ear power supply will only draw current from the AC mains when the rectified line voltage is below a fixed value. This condition will occur from a predetermined time before a line voltage zero crossing to a predetermined time after the line voltage zero crossing. The cut-out and cut-in voltage points can be adjusted so that the cat ear power supply  2810  draws current during a first interval from a time just after the line voltage zero crossing to a time when the bus capacitor  916  begins drawing charging current from the line, and during a second interval from a time when the bus capacitor  916  stops drawing charging current from the line until the next line voltage zero crossing.  
         [0089]     When the rectified line voltage is lower than a predetermined voltage, a charging transistor  2812  conducts to allow charging of energy-storage device capacitor  2814 , which charges toward a voltage VCC. The rate of charge of the capacitor  2814  is determined by a resistor  2816  in series with the drain of the MOSFET transistor  2812 , which, in conjunction with the value of the capacitor  2814 , can be chosen so as to substantially match the value of VCC to the predetermined cut-out and cut-in voltages. This allows for a substantially piece-wise continuous ballast input current. Although the transistor  2812  is shown as a MOSFET, it may be any suitable controllably conductive device, such as, without limitation, a BJT or an IGBT.  
         [0090]     When the rectified line voltage is equal to or greater than the predetermined voltage, then cut-out transistor  2818  begins conducting. The collector of the cut-out transistor  2818  pulls the cathode of a Zener diode  2820  toward VCC, which effectively turns off the charging transistor  2812 . The predetermined cut-in and cut-out voltages are determined by the resistive voltage divider network including resistors  2822  and  2824 , to which the base of the cut-out transistor  2818  is connected.  
         [0091]     In a second embodiment  2910  of the cat ear power supply  884 , shown in  FIG. 30 , the cat ear supply  2910  includes a circuit that actively monitors current drawn from the back end of the ballast and only draws current from the line when the back end is not drawing current above a predetermined value. The current monitor circuit includes transistor  2930 , capacitor  2932 , resistors  2934  and  2936 , and diodes  2938  and  2940 . When the ballast back end is drawing current above the predetermined value, the voltage at the emitter of transistor  2930  goes negative by a voltage equivalent to the combined forward voltage drops of diodes  2938 ,  2940 . The resulting forward bias across the base-emitter junction of transistor  2930  turns transistor  2930  on, effectively pulling the gate of transistor  2812  low, thereby turning off transistor  2812 . When back end current falls below the predetermined value, capacitor  2932  discharges to ground, transistor  2930  turns off, allowing transistor  2812  to turn on and provide a charging path for capacitor  2814 . This second embodiment yields a slight improvement in THD over the first embodiment.  
         [0092]     In addition to providing a means for shaping the input current drawn by the ballast so as to improve power factor and THD, the cat ear power supply advantageously provides a faster start-up of the ballast and is not affected by the operating mode of the ballast in the same way that typical prior art trickle-charge and bootstrap systems are affected.  
         [0093]     The result of combining the improved valley-fill circuits, control circuits, and cat ear power supply of the present invention may be seen in  FIG. 31 .  
         [0094]     The cat ear power supply comprises means for drawing input current near the zero crossing of the AC input voltage waveform so that the input current THD is substantially reduced. In other words, the cat ear power supply fills in the current waveform near the zero crossings.  
         [0095]     The improved valley-fill circuits of the invention comprise means for storing energy in an energy storage device over a substantial portion of each half-cycle of the AC input voltage so that the ballast power factor and THD are substantially improved. This is depicted in  FIG. 31  wherein it may be seen that the middle portion of each line half-cycle current waveform conforms substantially to an ideal sinusoidal current waveform.  
         [0096]     The combination of the cat ear power supply and the improved valley-fill circuits comprises means for selectively drawing current from the AC power supply so that the input current to the ballast is essentially sinusoidal.  
         [0097]     The operation of the ballast is further enhanced by the control circuit disclosed herein which comprises means for selectively varying the complementary duty cycles of the inverter switches in response to the bus voltage so as to cause an energy storage device to draw more current from the AC power supply around the peak of each line half-cycle of the bus voltage, and draw less current near the troughs of each line half-cycle of the bus voltage.  
         [0098]     While the embodiments heretofore described have been primarily directed to improving the THD, lamp current crest factor, and power factor of an electronic ballast, either of the dimming or non-dimming variety, the inventors believe that provision of an independent power supply in an electronic ballast is itself novel. That is, insofar as the inventors have been able to discover, all prior art electronic ballasts that have included control circuits for controlling the operation of an inverter stage have drawn the power for operating such control circuits from a transformer within the ballast, usually associated with the inverter stage, or an AFPC stage. This is so because such transformers have been present for the normal operating functions of the ballast. It is very easy to provide an additional secondary winding on such transformers for the purpose of drawing off power to drive the control circuits.  
         [0099]     Unfortunately, while it may be more cost effective to simply add another secondary winding to an already present transformer, it becomes more complicated to start up the ballast in a controlled manner upon application of power. This has lead to more complex circuitry for handling abnormal and transient conditions within such ballasts.  
         [0100]     The provision of an independent power supply, that is, one that derives its power directly from the line at the front end of the ballast, either through the ballast&#39;s own rectifier stage, or through its own dedicated rectifier, rather than from the secondary of a transformer associated with the ballast back end, greatly simplifies the handling of transient conditions upon start up, shut down, and during abnormal or fault conditions. In the present case, the preferred form of such an independent power supply is the previously described cat ear power supply. Thus, the independent power supply of the preferred embodiment allows one to decouple the power supply from the back end, thereby simplifying control of the ballast, while simultaneously providing a means to more precisely control the manner in which current is drawn from the line so as to reduce THD and improve power factor.  
         [0101]     Although the present invention has been described in relation to particular embodiments thereof, many other variations and modifications and other uses will become apparent to those skilled in the art. It is preferred, therefore, that the present invention be limited not by the specific disclosure herein, but only by the appended claims.