Abstract:
A control circuit is provided for regulating the rotational speed of a brushless DC motor by pulse width modulating at least one power transistor to pass a motor supply signal to the motor armature. A voltage averaging circuit generates an averaged signal indicative of the average voltage level being supplied to the motor. The averaged voltage signal is compared against a reference voltage to determine motor speed error in order to maintain the rotational speed of the motor at a generally constant level. A sawtooth or other periodic ramp signal is added to a motor current signal, and this composite signal is monitored by a comparator until it overcomes the motor speed error signal. The pwm circuit thereby modulates the power supply to regulate motor speed while maintaining a symmetrical motor armature current waveform. The control circuit takes advantage of the inherent inductance of the motor windings and the moment of inertia of the rotor assembly as filters to help smooth the physical operation of the motor and to further maintain its desired rotational speed.

Description:
CROSS-REFERENCE TO RELATED APPLICATION  
       [0001]    This patent application is a continuation-in-part of U.S. patent application Ser. No. 09/131,046, filed Aug. 7, 1998, co-pending herewith, and incorporated by reference herein. 
     
    
     
       FIELD OF THE INVENTION  
         [0002]    The present invention relates generally to a motor controller, and more particularly to a controller for controlling the rotational speed and armature current of a brushless DC motor.  
         BACKGROUND OF THE INVENTION  
         [0003]    Control circuits are known for controlling brushless DC motors, such as, for example, regulating the rotational speed of brushless DC fan motors that cool the interiors of computers. One problem with brushless DC fan motors is that they traditionally have had a narrow usable input range. Fan speed and input current are approximately proportional to input voltages. Thus, if the input voltage from an unregulated source such as a battery were used to power a brushless DC fan, such as a typical 24 volt nominal battery, the voltage would vary from about 28 volts in float state to about 21 volts in discharged state. This change would cause a brushless DC fan rated at a nominal 3500 RPM to vary as much as about 1000 RPM over the above-mentioned range of battery voltages. Such a large variation in RPM means that the fan is not properly cooling a computer at the low-end of the RPM range, and that power is being wasted at the high-end of the RPM range.  
           [0004]    Some brushless DC fan users have multiple input source voltages that their equipment is expected to operate from, with 24 volt and 48 volt systems being the most common. Such multiple source voltages pose the same problem in resultant RPM variation in a brushless DC fan motor as does a single input voltage source whose voltage level varies widely. Accordingly, there is a need to provide a brushless DC fan motor having a high input range with relatively little variation in motor rotational speed. For example, in the telecommunications industry, there is a need to provide a brushless DC fan motor having an input range of about 20-60 volts with little variation in motor rotational speed. However, other input voltage ranges may be provided for other motor applications.  
           [0005]    Linear regulators have been used to regulate brushless DC fan rotational speed. However, the linear regulator approach poses an efficiency problem. A brushless DC fan that draws 18 watts at 21 volts will draw almost 27 watts when operating at 28 volts, and 54 watts at 56 volts input, with the increase in power draw having to be dissipated as heat.  
           [0006]    Pulse width modulation (“pwm”) has also been used in the prior art to regulate motor speed. One method commonly used is to pulse width modulate the commutation transistors to the brushless DC motor. This employment of pulse width modulation reduces the dissipation of energy involved with changing motor speed. However, pulse width modulating the commutation transistors does not permit large changes in input voltage without widely varying the rotational speed of the brushless DC motor. This method is most commonly used in thermal brushless DC fans to reduce brushless DC fan speed at low temperatures. The speed variation is unfortunately even wider than that of the non-speed controlled type, and clamp dissipation is still relatively high.  
           [0007]    Another pwm approach is to use a fill bridge driver. This involves placing a bipolar motor winding between the legs of four switching transistors and controlling the timing of the pwm modulator and commutation logic to regulate motor current. Wide input voltage ranges are possible with high efficiency. A well designed full bridge driver can regulate motor speed over a better than 3:1 range of input voltage. The chief drawbacks are complicated logic and the difficulties of driving the four switching transistors without cross conduction through the series connected pairs. Although many manufacturers offer integrated full bridge devices, most suffer from a limitation of current and/or voltage.  
           [0008]    Another approach is to employ a pwm switching voltage regulator to accommodate a wide range of input voltages without widely varying the rotational speed of the motor. However, this requires relatively bulky filter inductors and capacitors.  
           [0009]    Of the above-mentioned pwm approaches, the pwm voltage regulator regulates motor voltage. The other methods typically regulate motor current. Voltage regulation is preferred to minimize variations in desired brushless DC motor speed. In other words, the variation in motor speed from motor to motor for a given current is greater than the variation in motor speed for a given voltage. Additionally, motor torque is a function of motor current.  
           [0010]    Therefore, if motor current is reduced in order to reduce motor speed to a low value, the motor torque becomes low. This means that the motor speed is sensitive to applied load (i.e., back pressure). This sensitivity to back pressure results in large speed deviations from the desired value. Motor-starting at low desired speeds is also a problem in that if the motor current is set too low then the motor will not be able to overcome the magnetic detents used to position the rotor away from the null point. Unfortunately, controlling motor voltage while failing to control motor current to adhere to a symmetrical waveform has the potential to increase vibration and electrical interference.  
           [0011]    Fans typically use one of two types of two-phase DC brushless motors, unipolar or bipolar. The difference between the two types is that a unipolar motor energizes two opposing poles of the four poles available, whereas a bipolar motor will energize all four poles at the same time, with the coils in quadrature having opposite magnetic polarity. Simply stated the unipolar type uses two pairs of coils with one pair energized and the other pair not energized, with the poles always energized in the same polarity. The bipolar motor energizes the four poles at the same time with adjacent poles having opposite polarity. Rotation of the motor of the unipolar type is accomplished by alternating energized pairs while the bipolar motor changes the polarity of the four poles.  
           [0012]    The bipolar motor has double the output of the unipolar motor because all of the copper is utilized and all four poles act upon the magnet. Drive complexity is greater as the direction of current must be reversed rather than just interrupted. In both cases however a problem of asymmetrical current in the motor exists. The current in the motor windings is reversed twice for each complete revolution of the bipolar motor. Various factors influence or modify the symmetry of the motor such as the degree of magnet strength, offset in the position sensor, mechanical variations in the motor components, and variations in wire resistance. This causes the current levels and the waveform shapes to differ from each other within a rotational period and allow different torques to be applied to the rotor, increasing vibration and noise. Accordingly, it would be desirable to provide an apparatus and method which may correct such non-ideal behavior in both unipolar and bipolar motors.  
           [0013]    It is also an object of the present invention to provide a brushless DC motor regulator which handles a relatively wide range of input voltages with little variation in the rotational speed of the motor.  
           [0014]    It is another object of the present invention to provide a brushless DC motor regulator which controls motor armature current to a substantially symmetrical waveform.  
           [0015]    It is a further object of the present invention to provide a brushless DC motor regulator that eliminates the relatively bulky filter capacitors and inductors interfacing the regulator and motor.  
           [0016]    The above and other objects and advantages of the present invention will become more readily apparent when the following detailed description is read in conjunction with the accompanying drawings.  
         SUMMARY OF THE INVENTION  
         [0017]    According to one aspect of the present invention a control circuit for controlling the rotational speed of a brushless DC motor is provided. The control circuit includes an electrical conduction switch having an input, an output, and a control terminal for passing a motor supply signal to a brushless DC motor from a voltage across first and second terminals of a DC voltage source. The input terminal of the switch is to be coupled to the first terminal of the DC voltage source, and the output terminal of the switch is to be coupled to the first terminal of the brushless DC motor. A voltage averaging circuit is provided having first and second input terminals and an output terminal for averaging the voltage level of the motor supply signal. The first input terminal of the voltage averaging circuit is coupled to the output of the switch, and the second terminal of the voltage averaging circuit is to be coupled to the second terminal of the voltage source. A differential amplifier has first and second input terminals and an output terminal for generating a signal corresponding to motor speed error. The first input terminal of the differential amplifier is coupled to a voltage reference potential indicative of the desired motor speed, and the second input terminal of the differential amplifier is coupled to the output terminal of the voltage averaging circuit.  
           [0018]    A pulse width modulator (“pwm”) of the invention has first and second input terminals and an output terminal. The first input terminal of the pwm is coupled to the output terminal of the differential amplifier for receiving the signal corresponding to motor speed error, the second input terminal of the pwm is coupled to a signal corresponding to the change in motor current, and the output terminal of the pwm is coupled to the control terminal of the electrical conduction switch. The pwm turns the switch on at a periodic rate, and turns the switch off after a delay, or pulse width, indicative of the difference in voltage level between the signal corresponding to motor speed error and the signal corresponding to change in motor current, in order to provide a motor supply signal having a substantially constant average voltage level corresponding with the desired motor speed and a substantially symmetrical current waveform. Preferably, the motor windings serve as an inductive filter to help smooth changes in current, and the rotor mass of the motor serves to help smooth the rotational speed of the motor.  
           [0019]    According to another aspect of the present invention, a control circuit for controlling the rotational speed of a brushless DC motor is provided. The control circuit includes first means to be coupled to an electrical power source for switchably passing a motor supply signal to a brushless DC motor. A second means is coupled to an output of the first means for generating an averaged signal by averaging the voltage of the motor supply signal. A third means is coupled to an output of the first means for generating a signal indicative of the change in motor current. A fourth means is coupled to an output of the second means for generating a speed error signal having a voltage level indicative of the difference in voltage between the voltage level of the averaged signal of the second means and a reference voltage. A fifth means turns on the first means periodically, and turns off the first means following a delay corresponding to the difference between the value of the speed error signal and the value of the change in motor current signal. These means provide a substantially constant average motor supply voltage level resulting in a substantially constant motor speed approximately equal to a desired motor speed, and a substantially symmetrical motor current supply signal waveform.  
           [0020]    According to yet another aspect of the present invention, a method of controlling the rotational speed of a brushless DC motor is provided. A motor supply signal is switchably passed from an electrical power source to a brushless DC motor. The voltage level of the motor supply signal is averaged to form an averaged signal. An error signal is generated having a voltage level indicative of the difference in voltage between the averaged signal and a reference voltage. A motor current signal is generated having a voltage level indicative of the change in current of the motor supply signal. The motor supply signal is modulated in response to the difference in value between the error signal and the motor current signal in order to provide a substantially constant voltage level and a substantially symmetrical motor current waveform.  
           [0021]    One advantage of the present invention is that the motor voltage signal is compared against the reference voltage to generate an error signal, and the error signal is in turn compared against the motor current signal to pulse width modulate the motor input signal. Accordingly, the apparatus and method of the present invention employ both a voltage feedback loop, and a current feedback loop embedded within the voltage feedback loop to maintain a substantially constant motor speed over a wide range of power supply voltages, to accurately select and control motor speed, and to do so while maintaining a substantially symmetrical armature current waveform. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0022]    [0022]FIG. 1 illustrates schematically a prior art electrical circuit of a pwm voltage regulator employing filter capacitors and inductors interfacing the regulator to a brushless DC motor.  
         [0023]    [0023]FIG. 2 illustrates schematically an electrical circuit of a pwm voltage regulator for a unipolar motor embodying the present invention which employs the brushless DC motor windings and rotor mass as a substitute for additional filter inductors and capacitors.  
         [0024]    [0024]FIG. 3 illustrates schematically a pwm sub-circuit of the pwm voltage regulator of FIG. 2.  
         [0025]    [0025]FIG. 4 illustrates schematically an alternative current compensating pwm sub-circuit embodying the present invention and which may form a part of the pwm voltage regulator of FIG. 2.  
         [0026]    FIGS.  5 A- 5 C illustrate three current waveform inputs to a motor demonstrating typical waveform improvements of the pwm sub-circuit of FIG. 4 when used in the pwm voltage regulator of FIG. 2.  
         [0027]    [0027]FIG. 6 illustrates schematically an electrical circuit of a pwm voltage regulator for a bipolar motor embodying the present invention and which also employs the brushless DC motor windings and rotor mass as a substitute for additional filter inductors and capacitors.  
         [0028]    [0028]FIG. 7 illustrates schematically a current compensating pwm control circuit embodying the present invention for the pwm voltage regulator of FIG. 6.  
         [0029]    FIGS.  8 A- 8 B illustrate two current waveform inputs to a motor operating at about 2000 RPM demonstrating typical waveform improvements of the pwm circuit of FIG. 7 when used with the pwm voltage regulator of FIG. 6.  
         [0030]    FIGS.  9 A- 9 B illustrate two current waveform inputs to a motor operating at about 3500 RPM demonstrating typical waveform improvements of the pwm circuit of FIG. 7 when used with the pwm voltage regulator of FIG. 6.  
         [0031]    FIGS.  10 A- 10 B illustrate two motor winding current waveforms in a motor operating at about 2000 RPM demonstrating typical waveform improvements of the pwm circuit of FIG. 7 when used with the pwm voltage regulator of FIG. 6.  
         [0032]    FIGS.  11 A- 11 B illustrate two motor winding current waveforms in a motor operating at about 3500 RPM demonstrating typical waveform improvements of the pwm circuit of FIG. 7 when used with the pwm voltage regulator of FIG. 6. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0033]    A prior art pwm voltage regulator will first be explained as background to the pwm voltage regulator embodying the present invention. Referring now to the prior art of FIG. 1, a pwm voltage regulator is indicated generally by the reference number  10  and is employed to control the rotational speed of a brushless DC motor  12  enclosed by dashed lines. The regulator  10  includes a positive terminal  11  and a negative terminal  13  for receiving a regulator DC input voltage from a power source (not shown). The pwm voltage regulator  10  includes an input filter  2 (capacitor  14 , a pwm power transistor  16  that is switched on and off by a pwm modulator  18 , a catch diode  20 , and an output filter circuit  22  enclosed by dashed lines which includes an output filter inductor  24  and an output filter capacitor  26 . The output filter inductor  24  and the output filter capacitor  26  are typically rather bulky, thereby imposing design constraints in relation to the increasing demand for smaller voltage regulators that are either separate from or incorporated in brushless DC motors. The demand for smaller regulated motors is particularly high in the computer industry which uses DC fan units incorporating regulated brushless DC motors for cooling electronic components.  
         [0034]    The output filter circuit  22  smoothes a pwm waveform generated by the pwm modulator  18  and the switching transistor  16  into a motor supply signal having an average DC voltage level. This average DC voltage level of the DC motor input signal determines the rotational speed of the motor  12 . In order to maintain the rotational speed of the brushless DC motor at a generally constant revolutions per minute (rpm), feedback is provided to the regulator  10 . To provide feedback, the voltage level of the DC motor input signal is received at the inverting input of a differential or error amplifier  28  and compared with a reference voltage V ref  which is provided at the non-inverting input of the amplifier  28 . The output of the error amplifier  28  is the difference between the two inputs to the error amplifier  28 , and is multiplied by the gain of the error amplifier. This error voltage output by the error amplifier  28  is provided as a feedback signal to the pwm modulator  18  which adjusts the width of the pulse or modulator signal generated by the pwm modulator. The modulator signal adjusts the switching on and off time of the transistor  16  in order to modulate the regulator input signal, which in turn adjusts the average voltage level of the motor input signal after being smoothed by the filter circuit  22 , in order to compensate for deviations in the motor input voltage level sensed by the feedback circuitry. The compensation thus tends to maintain the voltage level of the motor input signal constant despite changes in the voltage level of the regulator input signal or changes to the load in order that the rotational speed of the motor  12  remains relatively constant. As previously mentioned, a drawback with the prior art circuit is that the output filter inductor and capacitor  24 ,  26  are relatively bulky and therefore require considerable mounting space in the regulator circuitry. This large space requirement hampers the growing demand to incorporate brushless DC motors and regulator circuitry in ever smaller spaces, such as the relatively small spaces allotted for regulated DC cooling fan units within portable computers.  
         [0035]    Turning now to FIG. 2, a pwm voltage regulator circuit  100  is employed in a center tap modulation approach for regulating the rotational speed of a brushless DC motor  102  enclosed by dashed lines. The motor  102  is a conventional brushless DC motor which may be coupled to a fan  104  used to cool a surrounding area such as the inside of a computer. The motor  102  includes first and second directional windings  106 ,  108 , respectively. Each of the windings  106 ,  108  has a first end coupled to an input terminal or center tap  110  of the motor. A second end  112  of the first winding  106  is coupled to ground potential via a first commutation switching transistor  114 , and a second end  116  of the second winding  108  is likewise coupled to ground via a second commutation switching transistor  118 . The switching transistors  114 ,  118  are alternately turned on and off by means of a conventional commutator logic circuit  120 . A rotor  122  is caused to rotate, and in turn rotate the fan  104  coupled thereto, by interacting with an electromagnetic field generated by commutated current flowing through the first and second windings  106 , 108 .  
         [0036]    The voltage regulator circuit  100  includes a positive input terminal  124  and a negative input terminal  126  for receiving thereacross a DC regulator input signal from a power source (not shown). An input capacitor  128  is coupled across the positive and negative input terminals  124  and  126 . Means for switchably passing a motor supply signal to the brushless DC motor  102  includes, for example, a pwm power or switch or transistor  130 , such as an npn bipolar junction transistor (BJT). The transistor  130  has its collector  132  coupled to the positive input terminal  124  and its emitter  134  coupled to an input voltage terminal of the motor  102  at  110 . A catch diode  135  has its cathode coupled to the emitter  134  of the transistor  130  and its anode coupled to ground potential. Means for averaging the voltage of the motor supply signal includes a series connected resistor  136  and capacitor  138  which cooperate to form a voltage integrator. The resistor  136  and the capacitor  138  are coupled between the emitter  134  of the transistor  130  and the negative input terminal  126 . More specifically, the resistor  136  has respective first and second terminals  139 ,  140 , and the capacitor  138  has respective first and second terminals  141 ,  142 . The first terminal  139  of the resistor  136  is coupled to the emitter  134  of the transistor  130 . The second terminal  140  of the resistor  136  is coupled to the first terminal  141  of the capacitor  138  at a junction  144  where an averaged signal indicative of the average voltage level of the motor supply signal is generated, and the second terminal  142  of the capacitor  138  is coupled to the negative input terminal  126 .  
         [0037]    Means for generating a differential signal having a voltage level indicative of the difference between the voltage level of the averaged signal and a reference voltage includes a high gain operational or error amplifier  146 , such as a differential voltage amplifier. The error amplifier  146  has its inverting input coupled to the junction  144  via a resistor  147 . The gain of the error amplifier  146  is preferably selected so that only millivolts of difference between the inverting and non-inverting inputs will drive the amplifier output to its extreme. A resistor  149  is coupled between the inverting input of the error amplifier  146  and ground potential. The non-inverting input of the error amplifier  146 , which is fed the reference voltage, is coupled to a voltage V cc  via a resistor  151 . The non-inverting input is also coupled to ground via a series connected resistor  153  and potentiometer  155 . An output  148  of the error amplifier  146  is coupled to a pulse width modulator (pwm)  150  at an input  152 . The pwm  150  is a means for turning on and off the transistor  130  to modulate the motor supply signal so that the motor supply signal is maintained at a substantially constant voltage level and, in turn, the motor  102  is maintained at a substantially constant rotational speed. A roll-off capacitor  157  is coupled between the inverting input and the output  148  of the error amplifier  146 . An output  154  of the pwm  150  is coupled to a base  156  of the transistor  130 .  
         [0038]    One preferred embodiment of the pulse width modulator  150  is illustrated in FIG. 3. The pwm  150  includes an oscillator sub-circuit  200  and a driver sub-circuit  300  each enclosed by dashed lines. The oscillator  200  includes a comparator  202  having its non-inverting input coupled to a V cc  source at  204  via a resistor  206 . An output  208  of the comparator  202  is coupled to its non-inverting input via a resistor  210 . A resistor  212  is coupled between the non-inverting input of the comparator  202  and ground potential. The resistors  206 ,  210  and  212  are coupled to one another at a junction  214 . A timing resistor  216  is coupled between the output  208  of the comparator  202  and the inverting input of the comparator. A timing capacitor  218  is coupled between the inverting input of the comparator  202  and ground potential.  
         [0039]    The driver sub-circuit  300  includes a comparator  302  having its non-inverting input coupled to the inverting input of the comparator  202  of the oscillator sub-circuit  200 . The inverting input of the comparator  302  at terminal  304  receives the error voltage signal from the output  148  of the error amplifier  146  shown in FIG. 2. An output  306  of the comparator  302  is coupled to a base of a transistor  308  via a resistor  314 . The transistor  308 , which serves as a pwm driver transistor, is shown as an npn BJT, but may be an FET or other suitable transistor for driving the pulse width modulator. An emitter of the transistor  308  is coupled to ground potential, and a collector of the transistor  308  is coupled at its output  310  to the base  156  of the power transistor  130  shown in FIG. 2.  
         [0040]    Referring now to the operation of the pwm voltage regulator circuit  100  shown in FIG. 2, the regulator circuit  100  receives a DC regulator input signal across the positive and negative input terminals  124 ,  126  from a power source (not shown) such as a DC power supply or an AC source that is rectified into DC voltage. The DC regulator input signal is initially filtered by the input capacitor  128  to further smooth the input voltage signal and to remove any unwanted transient voltage fluctuations. The motor supply signal derived from the DC power supply is modulated by the combination of the pwm  150  and the transistor  130  to generate a motor supply signal having a predetermined average voltage level suitable for operating the motor  102  at a desired rotational speed. Because the rotational speed of the motor  102  is a function of the voltage level of the motor supply signal, it is important to maintain this voltage level at substantially the same value.  
         [0041]    The pwm  150  sends a modulator signal to the base  156  of the switching transistor  130  to modulate the motor supply signal. The modulated motor supply signal present at the emitter  134  of the transistor  130  is the signal used to regulate the rotational speed of the motor  102 . The voltage level of the modulated motor supply signal is averaged at the junction  144  by the series combination of the resistor  136  and the capacitor  138  to form an averaged signal, and is thus indicative of the average voltage level of the motor supply signal. The voltage level of the averaged signal is a function of the pulse width of the modulated motor supply signal.  
         [0042]    The voltage level of this averaged signal at the junction  144  is reduced by the resistors  147 ,  149 , and this reduced voltage level of the averaged signal is received at the inverting input of the error amplifier  146  and compared with a reference, such as V ref , to generate a differential or error signal at the output  148  of the error amplifier  146 . V ref  is determined by the resistors  151 ,  153  and adjusted by the potentiometer  155 . The reference voltage V ref  is a fixed voltage level which is compared with the reduced voltage level of the averaged signal to determine if there is any deviation in the difference between the voltage level of V ref  and the reduced voltage of the averaged signal representing the motor supply signal or voltage, and thus indicating a tendency for the rotational speed of the motor  102  to change or drift over changes in input voltage to the pwm voltage regulator circuit  100 . As an example, the voltage level of the averaged signal V m  is selected as 12.75 volts, V ref  is 0.25 volt and the resistors  147 ,  149  are selected to reduce the voltage level of the averaged signal by a factor of 50 in order that V m =((resistance of the resistor  147 /resistance of the resistor  149 )*V ref )+V ref =((50)*0.25)+0.25)=12.75 volts.  
         [0043]    If there is a deviation between the ideal voltage V ref  (i.e., 0.25 volt in this example) and that of the voltage level of the reduced averaged signal at the inverting input of the error amplifier  146 , the error amplifier  146 , in order to compensate for any change in the voltage level of the motor supply signal (i.e., a deviation from 12.75 volts in this example), will generate an error signal at the output  148  of the amplifier  146  having a voltage magnitude proportional to the difference between the voltage levels present at the inverting and non-inverting inputs of the amplifier  146 . When the reduced voltage level of the averaged signal drops slightly in relation to V ref  because of, for example, a load increase or input voltage drop, the voltage level of the amplifier signal generated at the output  148  of the amplifier  146  will increase slightly. The increased voltage level of the amplifier signal will then be fed to the input  152  of the pwm  150  to slightly increase the duration or pulse width of the modulator signal generated at the output  154  of the pwm  150 .  
         [0044]    The increased duration of the modulator signal is fed to the base  156  of the power transistor  130  to increase the pulse width or duration of the turn-on time of the transistor  130 . The increased turn-on time thus increases the pulse width of the modulated motor supply signal present at the emitter  134  of the transistor  130  which is fed to the input voltage terminal  110  of the brushless DC motor  102 . The increased duration or pulse width of the modulated motor supply signal raises the average voltage level of the motor supply signal, to compensate for the slight drop in the voltage level of the motor supply signal, thereby maintaining the rotational speed of the motor at a generally constant rpm. Conversely, if the reduced voltage level of the averaged signal increases slightly, the amplifier  146 , the pwm  150  and the transistor  130  cooperate in a fashion opposite to that just described to decrease the pulse width of the motor supply signal for decreasing the average voltage level of the motor supply signal. The roll-off capacitor  157  is coupled across the error amplifier  146  to prevent the output of the error amplifier from slewing to its limits in response to V m  changes by means of reducing the high frequency gain of the error amplifier  146  to the point that the pwm  150  can follow the error amplifier output. The catch diode  135  prevents the inductor current from decaying at a rapid rate and the voltage at the center tap  100  from falling below ground potential in order to maintain the average voltage at the center tap as the motor current is being commutated.  
         [0045]    The pwm voltage regulator circuit just described is known to substantially maintain the rotational speed of a brushless DC motor over a wide range of motor supply voltages while also maintaining a symmetrical current waveform. An example of motor rotational speed and current as a function of motor supply voltage is set forth in Table 1.  
                       TABLE 1                       Voltage   Speed (RPM)   Current (Amperes)                   19   3680   0.82       20   3180   0.86       21   3210   0.84       30   3230   0.61       48   3230   0.41       60   3230   0.35                  
 
         [0046]    As can be seen from Table 1, over a motor supply voltage range of 20 volts to 60 volts, the rotational speed of a brushless DC motor controlled by the regulator circuit of the present invention is maintained substantially constant (i.e., the rotational speed varies 1.5%) as compared with conventional motors. As also shown in the Table, the rotational speed of the motor shows no discernible fluctuation over a motor supply voltage range of 30 to 60 volts.  
         [0047]    An advantage of employing the above-described voltage mode control for a brushless DC motor is that the control permits a high torque for starting the motor and a narrow speed for distribution range for tightly regulating the rotational speed of the motor over a large range of motor supply voltages. A further advantage of applying center tap modulation is that this type of modulation possesses the superior linear transfer characteristics found in full bridge modulation without certain drawbacks of full bridge modulation, including: the complexity of logic and sequencing of transistor switches, the possibility of cross conduction, and in turn, short circuiting across the input source, difficulty in sensing average motor coil voltage, sensing continuous motor current, and the high parts count inherent in employing full bridge modulation.  
         [0048]    The operation of the pwm  150  of FIG. 2 will be explained more fully with reference to FIG. 3. Preferably, the resistors  206 ,  210  and  212  are selected to be of equal resistance. When the output of the comparator  202  is low, the junction  214  of the resistors  206 ,  210  and  212  is at {fraction (1/3)} V cc . When the output of the comparator  202  is high, the junction is at {fraction (2/3)} V cc . The timing capacitor  218  is charged and discharged between {fraction (1/3)} V cc  and {fraction (2/3)} V cc  by the timing resistor  216 . The frequency of oscillation is primarily a function of the capacitance level of the timing capacitor  218  and the resistance level of the timing resistor  216 , and the duty cycle is preferably about 50%. A ramp voltage generated by the timing resistor  216  and the timing capacitor  218  is applied to the non-inverting input of the comparator  302  of the driver sub-circuit  300 . When the collector of the driver transistor  308  is low, the pwm switch transistor  130 , shown in FIG. 2, is off or non-conducting, and the center tap  110  of the motor  102  is at 0 volts. When the collector of the driver transistor  308  is high, then the pwm switch transistor  130  is on or conducting, and the center tap  110  of the motor  102  is coupled to V in .  
         [0049]    As can be seen in FIG. 2, there is no filter circuit external of the motor  102  for smoothing the voltage level of the motor supply signal in order to maintain the rotational speed of the motor  102  at a generally constant rpm. The regulator  100  uses the windings  106 ,  108  of the motor  102  to integrate the pwm voltage and to function similarly to the filter inductor  24  of FIG. 1, and changes in current drawn by the motor  102  are smoothed (i.e., the rotational speed of the motor is maintained substantially constant) by the mass of the rotor  122  in a similar way as the filter capacitor  26  of FIG. 1 smoothes the voltage level of the motor input signal. In other words, the motor inductance is substituted for the filter inductor  24  of FIG. 1, and the rotor mass or inertia is substituted for the filter capacitor  26  of FIG. 1. Thus, the bulky filter inductor  24  and the filter capacitor  26  of FIG. 1 are eliminated in the embodiment of FIG. 2. As a result, the regulator  100  consumes considerably less space than did prior regulators using bulky filter inductors and capacitors. Further, a regulated DC motor or DC fan unit that includes the regulator circuit embodying the present invention also consumes less space because of the elimination of the additional bulky filter components. Accordingly, the regulator circuit  100 , or a DC motor or DC fan unit incorporating the regulator embodying the present invention overcomes the space constraints that are found in the prior DC motors or DC fan units using additional filter components.  
         [0050]    Turning to FIG. 4, another preferred embodiment of the pulse width modulator is indicated generally by the reference numeral  150 ′. The pwm  150 ′ includes a periodic sub-circuit  400  and a driver sub-circuit  500  each enclosed by dashed lines. The periodic sub-circuit  400  comprises a periodic signal generator in the form of an oscillator comprising a comparator  402  and related circuit components. The comparator  402  has its non-inverting input coupled to a V cc  source at  404  via a resistor  406 . An output  408  of the comparator  402  is coupled to its non-inverting input via a resistor  410 . A resistor  412  is coupled between the non-inverting input of the comparator  402  and ground potential. The resistors  406 ,  410  and  412  are coupled to one another at a junction  414 . A timing resistor  422  is coupled between the output  408  of the comparator  402  and the cathode end of a diode  420 . The anode end of the diode  420  is coupled to the inverting input of the comparator  402 . A timing resistor  416  is coupled between the output  408  and the inverting input of the comparator  402 . A timing capacitor  418  is coupled between the inverting input of the comparator  402  and ground potential. A resistor  424  is coupled between the inverting input of the comparator  402  and a summing junction  430 . As described in further detail below, the periodic signal generator transmits a ramp signal to the junction  432  defining a sawtooth waveform. However, as may be recognized by those skilled in the pertinent art based on the teachings herein, the periodic signal generator may generate any of numerous different periodic or ramp signals suitable for performing the functions described herein. Similarly, the periodic signal generator may take any of numerous different configurations which now or later become known to those skilled in the pertinent art for performing the functions of the periodic signal generator described herein.  
         [0051]    Means for receiving a motor current sample signal are provided by an input terminal  428 . The input terminal  428  may be coupled, for example, to the common source terminal of the motor MOSFETs of FIG. 2. However, as may be recognized by those skilled in the pertinent art based on the teachings herein, the input terminal  428  may be coupled to any of numerous other motor current sources for generating the motor current signal described herein. A resistor  426  is coupled between the input terminal  428  and the summing junction  430 . Accordingly, the summing junction  430  provides a signal indicative of the sum of the periodic or ramp signal received from junction  432  and the motor current signal received from the input terminal  428 .  
         [0052]    The driver sub-circuit  500  includes a comparator  502  having its non-inverting input coupled to the summing junction  430  of the periodic sub-circuit  400 . The inverting input of the comparator  502  receives at terminal  504  the error voltage signal from the output  148  of the error amplifier  146  of FIG. 2. An output  506  of the comparator  502  is coupled to a Reset input of an RS flip-flop  512 . The Set input of flip-flop  512  is coupled to the output  408  of the periodic sub-circuit  400 . The inverting output of the flip-flop  512  is coupled through a resistor  514  to the base of a transistor  508 . An emitter of the transistor  508  is coupled to ground potential, and a collector of the transistor  508  is coupled at its output  510  to the base  156  of the power transistor  130  of FIG. 2. The transistor  508 , which serves as a pwm driver transistor, is shown as an npn BJT, but may be an FET or other suitable transistor or other electrical conduction switch for driving the pulse width modulator of the invention. Similarly, as may be recognized by those skilled in the pertinent art based on the teachings herein, the flip-flop  512  may take the form of any of numerous binary state or like devices which now or later become known to those skilled in the pertinent art for performing the functions of the flip-flop described herein.  
         [0053]    The resistors  406 ,  410  and  412  may be selected to be of equal resistance. Accordingly, when the output of the comparator  402  is low, the junction  414  of the resistors  406 ,  410  and  412  is at {fraction (1/3)} V cc , When the output of the comparator  402  is high, the junction  414  is at {fraction (2/3)} V cc . The timing capacitor  418  is periodically charged from {fraction (1/3)} V cc  to {fraction (2/3)} V cc  by the timing resistor  416 . The timing capacitor  418  is periodically discharged from {fraction (2/3)} V cc  to {fraction (1/3)} V cc  by the timing resistor  422  through diode  420 . The frequency of oscillation is primarily a function of the capacitance level of the timing capacitor  418  and the resistance levels of the timing resistors  416  and  422 . Timing resistor  416  determines the charge period, and the equivalent resistance of parallel resistors  416  and  422  determines the discharge period. Accordingly, a ramp voltage generated by the timing resistor  416  and the timing capacitor  418  is applied to the junction  432  and, in turn, to the summing junction  430 . Input terminal  428  passes a motor current signal across resistor  426  to summing junction  430 . Thus, the resultant signal at the summing junction  430  is approximately equal to the sum of the ramp signal and the motor current signal, and the summed signal is coupled to the non-inverting input of the comparator  502  of the driver sub-circuit  500 . The output terminal  506  of the comparator  502  is coupled to the Reset input of flip-flop  512 , thereby causing the flip-flop  512  to Reset whenever the value of the motor current plus ramp from the summing junction  430  exceeds the value of the error signal from the input terminal  504  as applied to the inverting input of the comparator  502 . The Set input of the flip-flop  512 , on the other hand, is activated every time the output  408  of the comparator  402  goes low, thereby activating the inverted output of flip-flop  512  at the start of each ramp cycle coinciding with the ramp signal received at the summing junction  430  across the resistor  424 . Once activated, the inverted output of the flip-flop  512  drives the collector of the driver transistor  508  high. When the collector of the driver transistor  508  is low, the pwm switch transistor  130  of FIG. 2 is off or non-conducting, and the center tap  110  of the motor  102  is at approximately 0 volts. When the collector of the driver transistor  508  is high, then the pwm switch transistor  130  is on or conducting, and the center tap  110  of the motor  102  is coupled to V in  in FIG. 2. As may be recognized by those skilled in the pertinent art based on the teachings herein, the periodic signal generator may generate any of numerous different periodic or ramp signals suitable for performing the functions described herein.  
         [0054]    [0054]FIG. 5A depicts a typical prior art pwm voltage regulator motor current waveform resulting when motor commutation happens to be ideal. As can be seen, the waveform is symmetrical from pulse to pulse, but non-symmetrical within each pulse. More frequently however, prior art motors will exhibit non-ideal commutation with a resultant motor current waveform such as that depicted in FIG. 5B. The waveform of FIG. 5B is non-symmetrical from pulse to pulse in addition to being non-symmetrical within each pulse. One advantage of the present invention is that a symmetrical motor current waveform, such as that depicted in FIG. 5C, is attainable via application of the present invention to brushless DC motors such as those used in the prior art.  
         [0055]    In the operation of the apparatus and method of the invention, the motor is powered by the voltage pulses passing across the pwm switch transistor  130  of FIG. 2 only when the output  510  of the transistor  508  is activated. The transistor  508  is activated periodically when the output  408  of the comparator  402  goes low, such activations corresponding to the start of a periodic sawtooth or other ramp signal generated by the oscillator or other periodic signal generator. The output  510  is effectively deactivated whenever the sum of the ramp signal at  432  and the motor current signal at  428  exceed the value of the error voltage  504  corresponding to the difference between motor actual speed and desired speed. Thus, motor speed is primarily controlled by the circuit of FIG. 2, while motor current is primarily controlled by the sub-circuit  400  of FIG. 4. The result is accurate motor speed control accompanied by symmetrical motor current waveforms, such as depicted in FIG. 5C. A further advantage of the symmetrical motor current waveform of the present invention is that it may have attendant acoustical benefits when realized in a fan motor controller for lower inertia fan assemblies.  
         [0056]    Turning now to FIG. 6, a pwm voltage regulator circuit  600  embodying the invention is employed in an H-bridge modulation approach for regulating the rotational speed of a bipolar brushless DC motor winding  622 . The motor is a conventional brushless DC motor which may be coupled to a fan used to cool a surrounding area such as the inside of a computer.  
         [0057]    The voltage regulator circuit  600  includes a positive input terminal  624  and a negative input terminal  626  for receiving thereacross a DC regulator input signal from a power source (not shown). An input capacitor  628  is coupled across the positive and negative input terminals  624  and  626 . Means for switchably passing a motor supply signal to a brushless DC motor winding  622  includes, for example, a first pwm switch or transistor  630 , and a second pwm switch or transistor  631 , such as the N-channel MOSFETs illustrated. The transistor  630  has its drain coupled to the positive input terminal  624  via junction terminal  610 , its source coupled to an input voltage terminal of the motor winding  622  at terminal  684 , and its gate coupled to a first pwm driver terminal  656 . The transistor  631  has its drain coupled to the positive input terminal  624  via junction terminal  610 , its source coupled to an input voltage terminal of the motor winding  622  at terminal  686 , and its gate coupled to a second pwm driver terminal  657 .  
         [0058]    The motor winding  622  has a first end coupled to a first input terminal  684 , and a second end coupled to a second input terminal  686 . The first input terminal  684  is alternately connected to ground potential via a first commutation switching transistor  614 , and the second input terminal  686  is likewise alternately connected to ground potential via a second commutation switching transistor  618 . The switching transistors  614 ,  618  are alternately turned on and off by means of a Hall Effect sensor  660 . The Hall Effect sensor  660  is coupled to a first output terminal  662  and a second output terminal  664 . Output terminal  662  is coupled to the gate of a commutation transistor  614 , which has its drain coupled to motor input terminal  684 . Output terminal  664  is coupled to the gate of a commutation transistor  618 , which has its drain coupled to motor input terminal  686 . The source terminals of commutation transistors  614  and  618  are coupled together and then coupled to motor current sample terminal  682 . Terminal  682  is dropped across resistor  680  to negative input terminal  626 , which is coupled to ground potential. Negative V CC  potential is coupled to the anode of a diode  672 , the cathode of which is connected to driver power terminal  668 . Terminal  668  is coupled to bootstrap capacitor  676  which is then coupled to motor input terminal  684 . Negative V cc  potential is also coupled to the anode of a diode  674 , the cathode of which is connected to driver power terminal  670 . Terminal  670  is coupled to bootstrap capacitor  678  which is then coupled to motor input terminal  686 . The motor winding  622  is caused to rotate by interacting with an electromagnetic field generated by commutated current flowing therethrough.  
         [0059]    Turning to FIG. 7, a preferred embodiment of the pulse width modulator for a bipolar motor is indicated generally by the reference numeral  650 . The pwm  650  includes a voltage sub-circuit  760 , a periodic sub-circuit  770 , and a driver sub-circuit  780  each enclosed by dashed lines.  
         [0060]    The voltage sub-circuit  760  comprises means for averaging the voltage of the motor supply signal including series connected resistors  636  and  637 , and an averaging capacitor  638  which cooperate to form a voltage averaging circuit. The resistor  636  is coupled between motor winding terminal  684  and terminal  644 , and the resistor  637  is coupled between motor winding terminal  686  and junction  644 . The averaging capacitor  638  is then coupled between junction  644  and ground potential. A first voltage dividing resistor  647  is coupled between junction  644  and the inverting input of an error amplifier  646 . A second voltage dividing resistor  649  is coupled between the inverting input of the error amplifier  646  and ground potential. The non-inverting input to the error amplifier  646  is coupled to a reference voltage V REF . The output of the error amplifier  646  is coupled to a voltage error terminal  652 . A capacitor  657  is coupled between the terminal  652  and the inverting input to the error amplifier  646 . A resistor  820  is coupled between terminal  652  and terminal  804 . A resistor  822  is coupled between terminal  804  and ground.  
         [0061]    Means for generating a differential signal at terminal  652  having a voltage level indicative of the difference between the voltage level of the averaged signal and a reference voltage includes the high gain operational or error amplifier  646 , which in the preferred embodiment of the present invention is a differential voltage amplifier. The error amplifier  646  has its inverting input coupled to the junction  644  via a resistor  647 . The gain of the error amplifier  646  is preferably selected so that only millivolts of difference between the inverting and non-inverting inputs will drive the amplifier output to its extreme. A resistor  649  is coupled between the inverting input of the error amplifier  646  and ground potential. The non-inverting input of the error amplifier  646  is fed the reference voltage V REF . An output terminal  652  of the error amplifier  646  is coupled to the driver sub-circuit  780  at terminal  804 . The pwm  650  provides a means for turning on and off the pwm switching transistors  630  and  631  of FIG. 6 to modulate the motor supply signal so that the motor supply signal is maintained at a substantially constant voltage level and, in turn, the motor winding  622  is maintained at a substantially constant rotational speed. A roll-off capacitor  657  is coupled between the inverting input and the output of the error amplifier  646 .  
         [0062]    The periodic sub-circuit  770  comprises a periodic signal generator in the form of an oscillator comprising a comparator  702  and related circuit components. The comparator  702  has its non-inverting input coupled to a V cc  source at  704  via a resistor  706 . An output  708  of the comparator  702  is coupled to its non-inverting input via a resistor  710 . A resistor  712  is coupled between the non-inverting input of the comparator  702  and ground potential. The resistors  706 ,  710  and  712  are coupled to one another at a junction  714 . A timing resistor  722  is coupled between the output  708  of the comparator  702  and the cathode end of a diode  720 . The anode end of the diode  720  is coupled to the inverting input of the comparator  702 . A timing resistor  716  is coupled between the output  708  and the inverting input of the comparator  702 . A timing capacitor  718  is coupled between the inverting input of the comparator  702  and ground potential. A resistor  724  is coupled between the inverting input of the comparator  702  and a summing junction  730 . As described in further detail below, the periodic signal generator transmits a ramp signal to the junction  732  defining a sawtooth waveform. However, as may be recognized by those skilled in the pertinent art based on the teachings herein, the periodic signal generator may generate any of numerous different periodic or ramp signals suitable for performing the functions described herein. Similarly, the periodic signal generator may take any of numerous different configurations which now or later become known to those skilled in the pertinent art for performing the functions of the periodic signal generator described herein.  
         [0063]    Means for receiving a motor current sample signal are provided by an input terminal  682 . The input terminal  682  may be coupled, for example, to the source terminals of the commutation switch MOSFETs of FIG. 6. However, as may be recognized by those skilled in the pertinent art based on the teachings herein, the input terminal  682  for receiving the motor current input signal may be coupled to any of numerous other motor current sources for generating the motor current signal described herein. A resistor  726  is coupled between the input terminal  682  and the summing junction  730 . Accordingly, the summing junction  730  provides a signal indicative of the sum of the periodic or ramp signal received from junction  732  and the motor current signal received from the input terminal  682 .  
         [0064]    The driver sub-circuit  780  includes a comparator  802  having its inverting input coupled to the summing junction  730  of the periodic sub-circuit  770 . The non-inverting input of the comparator  802  receives at terminal  804  the error voltage signal from the output  652  of the error amplifier  646  of FIG. 6. An output  806  of the comparator  802  is coupled to a Reset input of an RS flip-flop  812 . The Set input of flip-flop  812  is coupled to the output  708  of the periodic sub-circuit  700 . The non-inverting output of the flip-flop  812  is coupled to junction  824 . NAND gate  826  receives inputs from junction  824  and Hall output terminal  664 , and has its output coupled through a resistor  814  to the base of a transistor  808 . An emitter of the transistor  808  is coupled to ground potential, and a collector of the transistor  808  is coupled at its output  810  to the input  656  of the power transistor  630  of FIG. 6. NAND gate  828  receives inputs from junction  824  and Hall output terminal  662 , and has its output coupled through a resistor  815  to the base of a transistor  809 . An emitter of the transistor  809  is coupled to ground potential, and a collector of the transistor  809  is coupled at its output  811  to the input  657  of the power transistor  630  of FIG. 6. The transistors  808  and  809 , which serve as pwm driver transistors, are shown as npn BJTs, but may be FETs or other suitable transistors or other electrical conduction switches for driving the pulse width modulators of the invention. Similarly, as may be recognized by those skilled in the pertinent art based on the teachings herein, the flip-flop  812  may take the form of any of numerous binary state or like devices which now or later become known to those skilled in the pertinent art for performing the functions of the flip-flop described herein.  
         [0065]    When the output of the comparator  702  is low, the junction  714  of the resistors  706 ,  710  and  712  may be at {fraction (1/3)} V cc . When the output of the comparator  702  is high, the junction  714  may be at {fraction (2/3)} V cc . The timing capacitor  718  is periodically charged, for example, from {fraction (1/3)} V cc  to {fraction (2/3)} V cc  by the timing resistor  716 . The timing capacitor  718  is periodically discharged from {fraction (2/3)} V cc  to {fraction (1/3)} V cc  by the timing resistor  722  through diode  720 . The frequency of oscillation is primarily a function of the capacitance level of the timing capacitor  718  and the resistance levels of the timing resistors  716  and  722 . Timing resistor  716  determines the charge period, and the equivalent resistance of parallel resistors  716  and  722  determines the discharge period. Accordingly, a ramp voltage generated by the timing resistor  716  and the timing capacitor  718  is applied to the junction  732  and, in turn, to the summing junction  730 . Input terminal  682  passes a motor current signal across resistor  726  to summing junction  730 . Thus, the resultant signal at the summing junction  730  is approximately equal to the sum of the ramp signal and the motor current signal, and the summed signal is coupled to the non-inverting input of the comparator  802  of the driver sub-circuit  780 . The output terminal  806  of the comparator  802  is coupled to the Reset input of the flip-flop  812 , thereby causing the flip-flop  812  to Reset whenever the value of the motor current plus ramp from the summing junction  730  exceeds the value of the error signal from the input terminal  804  as applied to the inverting input of the comparator  802 . The Set input of the flip-flop  812 , on the other hand, is activated every time the output  708  of the comparator  702  goes low, thereby activating the inverted output of flip-flop  812  at the start of each ramp cycle coinciding with the ramp signal received at the summing junction  730  across the resistor  724 . Once activated, the non-inverted output of the flip-flop  812  drives the collector of one of the driver transistors  808  or  809  high. When the collector of the driver transistor  808  is low, the pwm switch transistor  630  of FIG. 6 is conducting to low potential. When the collector of the driver transistor  808  is high, then the pwm switch transistor  630  is conducting from high potential. Likewise, when the collector of the driver transistor  809  is low, the pwm switch transistor  631  of FIG. 6 is conducting to low potential. When the collector of the driver transistor  809  is high, then the pwm switch transistor  631  is conducting from high potential.  
         [0066]    As may be recognized by those skilled in the pertinent art based on the teachings herein, the periodic signal generator may generate any of numerous different periodic or ramp signals suitable for performing the functions described herein.  
         [0067]    Referring now to the method of operation of the pwm voltage regulator circuit  600  shown in FIG. 6, the two lower MOSFETs  614  and  618  are energized in response to the Hall Effect Sensor  660 , which provides two outputs  662  and  664  that are out of phase with each other. As may be reconfigured by those skilled in the pertinent art based on the teachings herein, this function also could be performed by a single output Hall Switch with an inverting buffer, a Hall element with suitable amplifier, or some other type of position sensor such as an opto/electric sensor. The purpose being to activate the associated transistor switch. Here, outputs are shown that are used to gate the PWM switches  630  and  631 . Alternatively, the output of the position sensor could be used to gate the synchronous switching of lateral pairs for greater power efficiency.  
         [0068]    The direction of the current in the motor winding  622  is determined by the appropriate activation of opposing transistor pairs  614  and  631 , or  618  and  630 . For example, when PWM Switch  630  and Commutation Switch  618  are “ON”, then one might say that the motor winding is energized “+ to −.” When the opposite case occurs where PWM Switch  631  and Commutation Switch  614  are “ON”, the winding would be energized “− to +.” By repeating this sequence the motor is caused to rotate, and by the reversing of the current the motor is caused to be a bipolar motor.  
         [0069]    Motor current is dropped across the resistor  680  to produce a small voltage at terminal  682  that is used to detect the amplitude of the motor current. This voltage representing the motor current is to be used in the course of regulating the motor current during normal and fault conditions. The MOSFETs  614 ,  618 ,  630  and  631  are N-channel types but could, by appropriate circuit design, be a mixture of P-channel and N-Channel types or some other type of solid state devices such as bipolar transistors or IBGTs.  
         [0070]    For the power circuits to function properly in this embodiment a bias source of sufficient amplitude, about 10 volts higher than +Vin, must be provided to the upper side PWM switches  630  and  631 . This is accomplished with the use of bootstrap drivers comprising the bootstrap capacitors  676  and  678 , and the rectifier diodes  672  and  674 . When the PWM switch  630  or  631  is “OFF”, then that PWM switch Source lead which has a common connection with the bootstrap capacitor  676  or  678  is at approximately ground potential. The other end of the bootstrap capacitor is connected to −Vcc through the associated rectifier  672  or  674 . When the PWM switch  630  or  631  is “ON”, the PWM Switch Drain and Source are both at nearly Vin potential. The bootstrap capacitor  676  or  678  which was charged to Vcc is now at Vcc+Vin. This provides the necessary voltage to energize the PWM Switch  630  or  631 .  
         [0071]    With reference to FIG. 7, the oscillator is composed of the comparator  702 , the resistor divider network comprising resistors  706 ,  710  and  712 , and a time constant network. The values of the resistors  706 ,  710  and  712  determine the oscillator ramp levels at terminal  732  which for a 10-volt Vcc could be around 2 volts for a ramp voltage valley and around 7 volts for a ramp voltage peak.  
         [0072]    The relative ratios for the charging resistor  716  and the discharging resistor  722  could be approximately 50:1. Ramp signal frequency is then determined primarily by charging resistor  716  and timing capacitor  718  and could be in the area of 20 kHz. The ramp signal from terminal  732  is summed at summing junction  730  with the Motor Current Sample from terminal  682  and applied to the inverting input of the second comparator  802 .  
         [0073]    In order to accurately maintain motor speed over a variety of input voltages V IN  and to minimize the variation between production motors it is desirable to use a voltage control loop. A constant voltage drive is believed to maintain motor speed better than a constant current drive over different load conditions, such as back pressure variations or air density differences, because the motor is allowed to draw more current and therefore do more or less work as required. Constant voltage drive also results in less speed variation from motor to motor under equal loading conditions because variations from motor to motor are expressed as differences in input current for each individual motor. A well designed constant current drive can, however, result in less current variation within the rotational period of the motor.  
         [0074]    By imbedding a current control loop within a voltage control loop in accordance with the invention optimal characteristics can be obtained. In other words, the voltage loop maintains a more equal speed over varying load conditions and minimizes speed deviations between different motors, and the current loop maintains a substantially constant current within the rotational period.  
         [0075]    To accomplish the voltage control the present invention compares the motor voltage against a reference level. This is done by differential amplifier  646  that has the non-inverting terminal connected to a reference voltage. For the purpose of example that reference could be 0.25 volts. The motor voltage is derived with an averaging capacitor  638  and two resistors  636  and  637  attached to either end of the motor winding. The voltage on the capacitor  638  is equal to about {fraction (1/2)} of the actual motor voltage. The divider network comprised of resistors  647  and  649  then scales that averaged voltage to be equal to the same 0.25 volts on the inverting terminal of the differential amplifier at the desired real motor voltage level.  
         [0076]    The gain of the amplifier  646  is very high such that typically less than 1 mv of difference between inputs will result in several volts of change in the output. This output voltage, referred to as the Error voltage, is in turn applied to the non-inverting input of a comparator through divider resistors  820  and  822 . This is to scale the full output voltage of the operational amplifier ( 646 - 842 ) to about 0.3 volts for 7.5 volts on the output  806  of the amplifier  802 .  
         [0077]    The motor current sample at terminal  682 , which has a level of about 0.2 volts for full load operation, is summed at junction  730  with a portion of the ramp obtained from the oscillator and applied to the inverting input of the comparator  802 . This causes the output of the comparator to go “LOW” when the instantaneous value of the motor sample plus ramp exceeds the voltage level set by the Error amplifier  802 .  
         [0078]    The output of the oscillator comparator  702  is “LOW” when the oscillator timing capacitor  718  is discharged, and this is connected to the “Set” input of flip-flop  812  so as to “Set” the flip-flop at the end of each oscillator cycle. The output of the comparator  802 , which compares the current sample to the Error voltage, is connected to the Reset input of the flip-flop  812 . The “Q” output of the flip flop  812  will then be “HIGH” when the oscillator starts the timing cycle, and will go “LOW” when the current sample exceeds a level determined by the voltage error amplifier  802 .  
         [0079]    The output or the flip-flop  812  is then applied to the inputs of two NAND gates  826  and  828  that also receive inputs from the Hall Effect Switch  660 . The NAND gates route the flip flop output to the appropriate drive transistor  808 - 809  according to the motor position as sensed by the Hall Effect Switch  660 , which in turn enables the appropriate PWM MOSFET  630  or  631 . Accordingly, the comparator  802  controls the current in the motor by affecting interruption of current flow through the PWM MOSFETs  630  or  631 . This occurs when the current exceeds a level set by the Voltage Error amplifier  646 . The amplitude of the motor current is determined by comparing the motor voltage against a desired value but nevertheless will remain substantially constant over the rotation period.  
         [0080]    [0080]FIGS. 8A and 9A depict typical prior art pwm voltage regulator motor input current waveforms resulting at motor speeds of about 2000 and 3500 RPM respectively. As can be seen, the input waveforms are non-symmetrical within each pulse and non-symmetrical between cycles. One advantage of the present invention is that substantially symmetrical motor input current waveforms, such as those depicted in FIGS. 8B and 9B, are attainable via application of the present invention to brushless DC motors such as those used in the prior art.  
         [0081]    [0081]FIGS. 10A and 11A depict typical prior art pwm voltage regulator motor winding current waveforms resulting at motor speeds of about 2000 and 3500 RPM respectively. As can be seen, the motor winding waveforms are non-symmetrical within each pulse and non-symmetrical between cycles. Another advantage of the present invention is that symmetrical motor winding current waveforms, such as those depicted in FIGS. 10B and 11B, are attainable via application of the present invention to brushless DC motors such as those used in the prior art.  
         [0082]    Those skilled in the pertinent art may recognize, based on the teachings herein, that the aforementioned method of motor speed regulation may be applied to time-varying motor control with relative ease by application of a time-varying motor reference signal to the present invention. Accordingly, the method of the present invention may further comprise the steps of controlling a pulse width of the pulsed motor supply signal to control the average voltage level of the pulsed motor supply signal in response to a control input via the reference voltage corresponding to a desired change in the angular velocity of the motor.  
         [0083]    Accordingly, although this invention has been shown and described with respect to exemplary embodiments thereof, it should be understood by those skilled in the art that the foregoing and various other changes, omissions, and additions in the form and detail thereof may be made therein without departing from the spirit and scope of the invention.