Abstract:
One object is to provide a front-end module with a shared output terminal wherein an input impedance is readily matched and an insertion loss is suppressed. In accordance with one aspect, the front-end module  10  includes an input terminal, output terminals, a first filter circuit that passes signals in a first passband, a second filter circuit that passes signals in a second passband, a switch that is disposed between the input terminal and the first and second filter circuits and selectively connects the input terminal to the first and second filter circuits, and a matching circuit. The second filter circuit includes phase shifters.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This application is based on and claims the benefit of priority from Japanese Patent Application Serial No. 2011-289511 (filed on Dec. 28, 2011), the contents of which are hereby incorporated by reference in their entirety. 
       TECHNICAL FIELD 
       [0002]    The invention relates to a front-end module, and in particular, to a front-end module used for a multi-band communication device. 
       BACKGROUND 
       [0003]    Multi-band mobile phones capable of handling telephone calls and transmitting data by a plurality of communication methods have been widely used. These multi-band mobile phones are generally provided with a front-end module made by integrating a RF circuit composed of one or more high-frequency switches, filters, and amplifying elements into a single package. The front-end module divides a received multi-band signal containing superimposed signals of a plurality of frequency bands into signals of individual frequency bands, and outputs the divided signal to a subsequent circuit such as a receiver. 
         [0004]    There are some publications proposing a downsized font-end module in which an amplifying element is shared by a plurality of frequency bands. For example, Japanese Patent Application Publication No. 2005-64778 (the “&#39;778 Publication”) discloses a front-end module with shared output terminals, wherein a pair of bandpass filters having different passbands are disposed in parallel between a common input terminal and common output terminals, and the input end of each of the pair of bandpass filters is connected to a switch and the output end of each of the pair of bandpass filters is connected to another switch (see  FIG. 8-B  in the &#39;778 Publication). 
       SUMMARY 
       [0005]    Thus, the front-end module disclosed in the &#39;778 Publication is provided with a switch at each of the input and output ends of the filters so that each of the filters is actuated while being separated from the other. Accordingly, the input impedance of each of the filters is readily matched to a reference impedance. However, this front-end module is subject to a large insertion loss because a received signal is attenuated when it passes through the switches. In view of these limitations, various embodiments of the present invention provide front-end modules with a shared output terminal wherein an input impedance is readily matched and an insertion loss is suppressed. Other objects of the present invention will be understood based on the following detailed description and attached drawings. 
         [0006]    A font-end module according to an embodiment of the present invention comprises: an input terminal; one or more output terminals; a first filter circuit comprising a first input port and one or more first output ports and disposed between the input terminal and the one or more output terminals; a second filter circuit comprising a second input port and one or more second output ports and disposed between the input terminal and the one or more output terminals; a switch disposed between the input terminal and the first and second input ports, for connecting the input terminal selectively to the first and second filter circuits; and a matching circuit disposed between the one or more output terminals and one or more junctions between the one or more first output ports and the one or more second output ports. In the embodiment, the first filter circuit comprises a first filter element including a third input port and one or more third output ports, for passing a signal in a first passband; and the second filter circuit comprises a second filter element including a fourth input port and one or more fourth output ports, for passing a signal in a second passband different from the first passband, and one or more first phase shifters disposed between the one or more fourth output ports of the second filter element and the matching circuit. In the embodiment, a phase of a reflection coefficient in the first passband of the second filter element as seen from the one or more fourth output ports under a condition where the switch is switched to disconnect the input terminal from the second filter circuit, the one or more first phase shifters are removed, and the one or more junctions are opened, is leading, by 0° to 180°, a phase of a reflection coefficient in the second passband of the first filter element as seen from the one or more third output ports under a condition where the switch is switched to disconnect the input terminal from the first filter circuit and the one or more junctions are opened. 
         [0007]    Thus, the various embodiments of the present invention provide front-end modules with a shared output terminal wherein an impedance is readily matched and an insertion loss is suppressed. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0008]      FIG. 1  is a circuit diagram of a front-end module according to an embodiment of the present invention. 
           [0009]      FIG. 2  is a Smith chart showing the frequency characteristics of an input impedance of a bandpass filter  46  as seen from balanced ports P 3 - 2 , P 3 - 3 . 
           [0010]      FIG. 3  is a Smith chart showing the frequency characteristics of an input impedance of a bandpass filter  48  as seen from balanced ports P 4 - 2 , P 4 - 3 . 
           [0011]      FIG. 4  is a Smith chart showing the frequency characteristics of an input impedance of a matching circuit  38  as seen from output terminals  40 . 
           [0012]      FIG. 5  is a Smith chart showing the frequency characteristics of an input impedance of a second filter circuit  36  as seen from junctions J 1 , J 2 . 
           [0013]      FIGS. 6   a  to  6   d  are schematic diagrams of phase difference between reflection coefficients. 
           [0014]      FIG. 7   a  is a Smith chart showing the frequency characteristics of an input impedance of the front-end module  10  (freed of phase shifters  50  and  52 ) as seen from an input terminal  12  and the output terminals  40  during operation of the bandpass filter  46 . 
           [0015]      FIG. 7   b  is a Smith chart showing the frequency characteristics of an input impedance of the front-end module  10  (including phase shifters  50  and  52 ) as seen from the input terminal  12  and the output terminals  40  during operation of the bandpass filter  46 . 
           [0016]      FIG. 8   a  is a Smith chart showing the frequency characteristics of an input impedance of the front-end module  10  (freed of phase shifters  50  and  52 ) as seen from an input terminal  12  and the output terminals  40  during operation of the bandpass filter  48 . 
           [0017]      FIG. 8   b  is a Smith chart showing the frequency characteristics of an input impedance of the front-end module  10  (including phase shifters  50  and  52 ) as seen from the input terminal  12  and the output terminals  40  during operation of the bandpass filter  48 . 
       
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       [0018]    Various embodiments of the present invention will now be described with reference to the attached drawings.  FIG. 1  is a circuit diagram showing a front-end module  10  according to an embodiment of the present invention. As shown, the front-end module  10  according to the embodiment comprises: a switch  14  connected to an antenna terminal  12 ; a first filter circuit  34  and a second filter circuit  36  each connected subsequently to the switch  14  via a switch-adapted matching circuit  32 ; a matching circuit  38  connected to the outputs of the first filter circuit  34  and the second filter circuit  36 ; and balanced output terminals  40 - a,    40 - b.  A multiband signal is inputted from an antenna (not shown) via the antenna terminal  12  and is transmitted selectively to one of the first filter circuit  34  and the second filter circuit  36  in accordance with the switching operation of the switch  14 . The signal is passed through the first filter circuit  34  or the second filter circuit  36  and is outputted to a subsequent receiver (not shown) from the output terminals  40 - a,    40 - b.  This front-end module  10  can also be used to send signals. That is, a transmitter (not shown) can be connected subsequently to the output terminals  40 - a,    40 - b,  and a signal from this transmitter can be wirelessly transmitted from an antenna via the antenna terminal  12 . In this specification, the output terminals  40 - a,    40 - b  are collectively termed “output terminals  40 ” or simply “output terminals.” 
         [0019]    The switch  14  in an embodiment is, for example, a single pole dual throw (SPDT) switch comprising a first terminal  16  connected to the antenna terminal  12 , a second terminal  18 , and a third terminal  20 . The switch  14  further comprises a voltage supplying terminal (not shown) for supplying voltage and a control terminal (not shown) for inputting a control signal. The first terminal  16  is connected selectively to one of the second terminal  18  and the third terminal  20  based on a control signal inputted from the control terminal. The switch  14  may have a desired number of terminals depending on a particular circuit configuration. For example, the switch  14  may be a SP3T switch, a SP4T switch, a SP8T switch, a DPDT switch, or a DP4T switch. An example of a switching element constituting the switch  14  is a field-effect transistor. 
         [0020]    Disposed subsequently to the switch  14  is the switch-adapted matching circuit  32  for matching an input impedance of the switch  14  to a reference impedance. The switch-adapted matching circuit  32  includes, for example, an inductor  42  of which one end is connected to the second terminal  18  of the switch  14  and the other end is connected to a first bandpass filter  46  (described later), and an inductor  44  of which one end is connected to the third terminal  20  of the switch  14  and the other end is coupled to a second bandpass filter  48  (described later). The input impedance of the switch  14  can be matched to the reference impedance by a publicly known method of adjusting the inductance values of the inductor  42  and the inductor  44 . The specific configuration of the switch-adapted matching circuit  32  is not limited to that illustrated in  FIG. 1  and may be modified appropriately. 
         [0021]    Disposed subsequently to the switch-adapted matching circuit  32  are the first filter circuit  34  and the second filter circuit  36 . The first filter circuit  34  comprises an unbalanced port P 1 - 1  and a pair of balanced ports P 1 - 2 , P 1 - 3 , and is connected to the switch-adapted matching circuit  32  via the unbalanced port P 1 - 1 . The second filter circuit  36  comprises an unbalanced port P 2 - 1  and a pair of balanced ports P 2 - 2 , P 2 - 3 , and is connected to the switch-adapted matching circuit  32  via the unbalanced port P 2 - 1 . The junction J 1  between the balanced port P 1 - 2  of the first filter circuit  34  and the balanced port P 2 - 3  of the second filter circuit  36  is connected to the output terminal  40 -a via the matching circuit  38 . The junction J 2  between the balanced port P 1 - 3  of the first filter circuit  34  and the balanced port P 2 - 2  of the second filter circuit  36  is connected to the output terminal  40 - b  via the matching circuit  38 . In this specification, the balanced port P 1 - 2  and the balanced port P 1 - 3  are also collectively termed “first output ports,” and the balanced port P 2 - 2  and the balanced port P 2 - 3  are also collectively termed “second output ports.” 
         [0022]    The first filter circuit  34  comprises the first bandpass filter  46  having a first passband. The first bandpass filter  46  transmits inputted signals within the first passband to the subsequent circuit while suppressing signals outside the first passband.  FIG. 1  shows, as an example of the first bandpass filter  46 , a balanced bandpass filter comprising an unbalanced port P 3 - 1  and a pair of balanced ports P 3 - 2 , P 3 - 3 . The balanced first bandpass filter  46  converts an unbalanced signal inputted from the unbalanced port P 3 - 1  into a balanced signal to be outputted from the balanced ports P 3 - 2 , P 3 - 3 . 
         [0023]    The second filter circuit  36  comprises the second bandpass filter  48  having a second passband, and the phase shifters  50 ,  52 . As the first bandpass filter  46 , the second bandpass filter  48  illustrated in  FIG. 1  comprises an unbalanced port P 4 - 1  and a pair of balanced ports P 4 - 2 , P 4 - 3 , wherein an unbalanced signal inputted from the unbalanced port P 4 - 1  is converted into a balanced signal and is outputted from each of the balanced ports P 4 - 2 , P 4 - 3 . In this specification, the balanced ports P 3 - 2 , P 3 - 3  are collectively termed “third output ports,” and the balanced ports P 4 - 2 , P 4 - 3  are collectively termed “fourth output ports.” The balanced port P 4 - 2  is connected to the phase shifter  50 , and the balanced port P 4 - 3  is coupled to the phase shifter  52 . The phase shifters  50 ,  52  comprise any distributed constant line such as a strip line or a micro strip line, and rotate the phase of an input signal by an amount in accordance with their line length. These phase shifters may consist of or comprise a concentrated constant element. In an embodiment, the wiring between the second bandpass filter  48  and the matching circuit  38  can be used as phase shifters  50 ,  52 . 
         [0024]    As will be described below, the phase of a reflection coefficient in the first passband of the second filter element  48  as seen from the balanced ports P 4 - 2 , P 4 - 3  (the fourth output ports) under the condition where the switch  14  is switched to disconnect the input terminal  12  from the second filter circuit  36 , the phase shifters  50 ,  52  are removed, and the junctions J 1 , J 2  are opened (i.e., the junctions J 1 , J 2  are disconnected from the filter element  48 ) leads, by 0° to 180°, the phase of a reflection coefficient in the second passband of the first filter element  46  as seen from the balanced ports P 3 - 2 , P 3 - 3  (the third output ports) under the condition where the switch  14  is switched to disconnect the input terminal  12  from the first filter circuit  34  and the junctions J 1 , J 2  are opened (i.e., the junctions J 1 , J 2  are disconnected from the filter element  46 ). In an embodiment of the present invention, the phase shifters  50 ,  52  are provided to a filter circuit (the second filter circuit  36  in  FIG. 1 ) including a bandpass filter (the second bandpass filter  48  in  FIG. 1 ) that generates a reflected wave of one of the first filter circuit  34  and the second filter circuit  36  that has a leading phase. 
         [0025]    In an embodiment, the first bandpass filter  46  and the second bandpass filter  48  may include a surface acoustic wave filter (SAW filter) and a bulk acoustic wave filter (BAW filter). The passbands of the first bandpass filter  46  and the second bandpass filter  48  may be, for example, various receiving bands or sending bands defined by Universal Mobile Telecommunication System (UMTS). The first bandpass filter  46  and the second bandpass filter  48  may be unbalanced bandpass filters that output unbalanced signals. 
         [0026]    The matching circuit  38  is configured such that the input impedance of the front-end module  10  as seen from the antenna terminal  12  and/or the output terminals  40  is matched to an external circuit connected subsequently to the output terminals  40  in the passbands of the first bandpass filter  46  and the second bandpass filter  48 . That is, the matching circuit  38  may be configured such that, when the first terminal  16  of the switch  14  is connected to the second terminal  18  to actuate the first bandpass filter  46 , the input impedance of the front-end module  10  is matched to a reference impedance in the passband of the first bandpass filter  46 , and when the first terminal  16  of the switch  14  is connected to the third terminal  20  to actuate the second bandpass filter  48 , the input impedance of the front-end module  10  is matched to a reference impedance in the passband of the second bandpass filter  48 . In an embodiment, the matching circuit  38  comprises an inductor  54  connected to the output terminals  40 - a,    40 - b.  The matching circuit  38  will be further described below. 
         [0027]    Thus, in a front-end module  10  according to an embodiment of the present invention, the output ports of the first filter circuit  34  and the second filter circuit  36  are connected to common output terminals  40 - a,    40 - b  via the matching circuit  38 , thereby suppressing an insertion loss compared to a conventional front-end module wherein a switch is provided subsequently to a pair of bandpass filters (see  FIG. 8-B  of The &#39;778 Publication). 
         [0028]    In a front-end module  10  according to an embodiment of the present invention, a matching circuit  38  (not a switch) is provided at the outputs of the first filter circuit  34  and the second filter circuit  36 , and thus the first filter circuit  34  and the second filter circuit  36  are constantly connected to each other. Accordingly, when the switch  14  is switched to actuate one of the first bandpass filter  46  and the second bandpass filter  48 , the non-actuated filter circuit acts as a reactance element on the actuated filter circuit. This action possibly causes the input impedance in the passband of the actuated filter circuit to deviate from the reference impedance, resulting in a deteriorated insertion loss of the front-end module  10 . For example, when the switch  14  is switched to connect the first terminal  16  to the second terminal  18  and to actuate the first bandpass filter  46 , the input impedance of the front-end module  10  is possibly deviated from the reference impedance in the first passband by the effect of the non-actuated second bandpass filter  48 . In order to address the limitations, the matching circuit  38  is configured such that the effect of the non-actuated filter circuit on the impedance matching of the actuated filter circuit is canceled (more specifically, the element values and arrangement of the reactance elements constituting the matching circuit  38  is adjusted). 
         [0029]    Each of the first bandpass filter  46  and the second bandpass filter  48  differently affects the impedance matching in the passband of the other (that is, the effect of the first bandpass filter  46  on the impedance matching in the second passband is different from the effect of the second bandpass filter  48  on the impedance matching in the first passband). As such, it is difficult to accurately cancel both the mutual effects of the first bandpass filter  46  and the second bandpass filter  48  on the impedance matching by only adjusting the element values of the reactance elements constituting the matching circuit  38 . In an embodiment of the present invention, the phase of a reflection coefficient in the second passband of the first bandpass filter  46  as seen from the third output ports (the balanced ports P 3 - 2 , P 3 - 3 ) under the condition where the switch  14  is switched to disconnect the first terminal  16  from the second terminal  18  (connect the first terminal  16  of the switch  14  to the third terminal  20 ) and the junctions J 1 , J 2  are opened, is compared to the phase of a reflection coefficient in the first passband of the second bandpass filter  48  as seen from the fourth output ports (the balanced ports P 4 - 2 , P 4 - 3 ) under the condition where the switch  14  is switched to disconnect the first terminal  16  from the third terminal  20  (connect the first terminal  16  of the switch  14  to the second terminal  18 ), the junctions J 1 , J 2  are opened, and the phase shifters  50 ,  52  are removed. Based on the comparison, it is determined that one or more phase shifters (the phase shifters  50 ,  52  in the example shown in  FIG. 1 ) are provided subsequently to the filter of which the phase of the reflection coefficient is leading by 0° to 180°. As will be described with reference to  FIGS. 2 to 5 , this arrangement accurately cancels the mutual effects of the first bandpass filter  46  and the second bandpass filter  48  on the impedance matching. Thus, the matching circuit  38  maintains the impedance matching in both the first bandpass filter  46  and the second bandpass filter  48 . 
         [0030]    Now with reference to  FIGS. 2 to 5 , an impedance matching using the phase shifters  50 ,  52  and the matching circuit  38  will be described.  FIG. 2  is a Smith chart showing the frequency characteristics of an input impedance of the first bandpass filter  46  as seen from the balanced ports P 3 - 2 , P 3 - 3  under the condition where the first terminal  16  of the switch  14  is disconnected from the second terminal  18  (the first terminal  16  of the switch  14  is connected to the third terminal  20 ) and the junctions J 1 , J 2  are opened.  FIG. 2  represents the simulation of an input impedance of the first bandpass filter  46  under the condition where a transformer is provided between the balanced ports P 3 - 2 , P 3 - 3  to render the first bandpass filter  46  single-ended.  FIG. 3  is a Smith chart showing the frequency characteristics of an input impedance of the second bandpass filter  48  as seen from the balanced ports P 4 - 2 , P 4 - 3  under the condition where the first terminal  16  of the switch  14  is disconnected from the second terminal  20  (the first terminal  16  of the switch  14  is connected to the second terminal  18 ), the junctions J 1 , J 2  are opened, and the phase shifters  50 ,  52  are removed. As does  FIG. 2 ,  FIG. 3  represents the simulation under the condition where the second bandpass filter  48  is rendered single-ended.  FIG. 4  is a Smith chart showing the frequency characteristics of an input impedance of the matching circuit  38  alone (i.e., an input impedance of the matching circuit  38  under the condition where the junctions J 1 , J 2  are opened in  FIG. 1 ) as seen from the output terminals  40 - a,    40 - b.  Further,  FIG. 5  is a Smith chart showing the frequency characteristics of an input impedance of the second filter circuit  36  as seen from the balanced ports P 2 - 2 , P 2 - 3  (second output ports) under the condition where the first terminal  16  of the switch  14  is disconnected from the third terminal  20  (the first terminal  16  of the switch  14  is connected to the second terminal  18 ), and the junctions J 1 , J 2  are opened. 
         [0031]    In the simulations, the passband of the bandpass filter  46  was set in the range of 925 to 960 MHz assigned for reception of the band VIII of UMTS, and the passband of the bandpass filter  48  was set in the range of 869 to 894 MHz assigned for reception of the band V of UMTS. The reference impedance on the unbalanced side was set at 50Ω, and the reference impedance on the balanced side was set at 100Ω. Further, both the inductance values of the inductors  42 ,  44  were set at 1.5 nH, and the inductance value of the inductor  54  was set at 13.5 nH. The line length for the phase shifters  50 ,  52  was set such that the phase shifters rotate an input signal of 0.95 GHz by 25°. In  FIGS. 2 to 5 , the central frequency of the passband of the first bandpass filter  46  is denoted by a marker m 2 , and the central frequency of the passband of the bandpass filter  48  is denoted by a marker m 1 . 
         [0032]    As shown in  FIG. 2 , the first bandpass filter  46  lies in a capacitive area in the Smith chart wherein the marker m 1  denotes the input impedance of the first bandpass filter  46 ; and thus the first bandpass filter  46  acts as a capacitive element on the second bandpass filter  48 . Accordingly, when the second bandpass filter  48  is actuated with the first terminal  16  of the switch  14  connected to the third terminal  20 , a signal passing through the second bandpass filter  48  is affected by the first bandpass filter  46  acting as a capacitive element. According to the simulation shown in  FIG. 2 , the imaginary component of the input impedance at the central frequency of the band V of the first bandpass filter  46  is about −74Ω. This effect of the first bandpass filter  46  on the impedance matching is canceled by the matching circuit  38 . That is, as shown in  FIG. 4 , the marker m 1  lies in an inductive area in the Smith chart showing the input impedance of the matching circuit  38 ; and thus, when the second bandpass filter  48  is actuated, a signal passing through the second bandpass filter  48  is affected by the matching circuit  38  acting as an inductive element. According to the simulation shown in  FIG. 4 , the imaginary component of the input impedance at the central frequency of the band V of the matching circuit  38  is about +74Ω. Thus, the matching circuit  38  is configured as follows: the imaginary component of an input impedance in the second passband of the first filter circuit  34  (which is equal to the input impedance in the second passband of the first bandpass filter  46  as seen from the balanced ports P 3 - 2 , P 3 - 3  in the example shown in  FIG. 1 ) as seen from the balanced ports P 1 - 2 , P 1 - 3  under the condition where the first terminal  16  of the switch  14  is disconnected from the second terminal  18  and the junctions J 1 , J 2  are opened, has an opposite polarity (the sign of positive or negative) than, and substantially the same amplitude as, the imaginary component of the input impedance in the second passband of the matching circuit  38  alone as seen from the output terminals  40 - a,    40 - b.  Accordingly, the matching circuit  38  can cancel the effect of the first bandpass filter  46  on the impedance matching occurring when the second bandpass filter  48  is being actuated. In this simulation, the effect on a passing signal in the second passband of the first bandpass filter  46  can be canceled by setting the inductance value of the inductor  54  at 13.5 nH. 
         [0033]    Meanwhile, as shown in  FIG. 3 , the marker m 2  lies in a capacitive area in the Smith chart showing the input impedance of the second bandpass filter  48 ; and thus the bandpass filter  48  acts as a capacitive element on the first bandpass filter  46 . Accordingly, when the first bandpass filter  46  is actuated with the first terminal  16  of the switch  14  connected to the second terminal  18 , a signal passing through the first bandpass filter  46  is affected by the second bandpass filter  48  acting as a capacitive element. According to the simulation shown in  FIG. 3 , the imaginary component of the input impedance at the central frequency of the band VIII of the second bandpass filter  48  is about −198Ω. Further, as shown in  FIG. 4 , the marker m 2  lies in the inductive area in the Smith chart showing the input impedance of the matching circuit  38 ; and thus the imaginary component of the input impedance of the matching circuit  38  at the central frequency of the band VIII is about 80Ω. Accordingly, the effect of the second bandpass filter  48  on the impedance matching occurring when the first bandpass filter  46  is being actuated cannot be canceled by only the matching circuit  38  optimized to cancel the effect of the first bandpass filter  46 . 
         [0034]    In an embodiment according to the present invention, the phase shifters  50 ,  52  provided subsequently to the second bandpass filter  48  varies the effect of the second bandpass filter  48  on a signal passing through the first bandpass filter  46 , and the matching circuit  38  also cancels the effect of the second bandpass filter  48 . In this simulation, the characteristic impedance of the phase shifters  50 ,  52  is set at 50Ω and the line length thereof is set such that the phase shifters rotate an input signal of 0.95 GHz by 25°. Thus, as shown in  FIG. 5 , the frequency characteristics of the input impedance of the second filter circuit  36  is equal to the frequency characteristics of the input impedance of the second bandpass filter  48  alone as shown in  FIG. 3  rotated clockwise by an angle in accordance with the line length of the phase shifters  50 ,  52 . In a Smith chart shown in  FIG. 5 , the marker m 2  lies at a position with a phase lag of the electrical length of the phase shifters (a position reached by rotating clockwise), as compared to the frequency characteristics of the input impedance of the second bandpass filter  48  alone shown in  FIG. 3 . Due to this phase rotation, the imaginary component of the input impedance at the central frequency of the band VIII of the second filter circuit  36  is about −80Ω. As stated above, the imaginary component of the input impedance at the central frequency of the band VIII of the matching circuit  38  is about +80Ω. Therefore, the effect of the second bandpass filter  48  on the impedance matching in the first passband occurring when the first bandpass filter  46  is being actuated can be canceled by the matching circuit  38 . 
         [0035]    Thus, the matching circuit  38  is configured as follows: the imaginary component of an input impedance in the second passband of the first filter circuit  34  as seen from the balanced ports P 1 - 2 , P 1 - 3  under the condition where the first terminal  16  of the switch  14  is disconnected from the second terminal  18  and the junctions J 1 , J 2  are opened, has an opposite polarity than, and substantially the same amplitude as, the imaginary component of an input impedance in the second passband of the matching circuit  38  alone as seen from the output terminals  40 - a,    40 - b.  Accordingly, the effect of the first bandpass filter  46  on the impedance matching occurring when the second bandpass filter  48  is being actuated can be canceled by the matching circuit  38 ; and the impedance matching in the second passband can be maintained. Accordingly, the phase shifters  50 ,  52  are configured as follows: the imaginary component of an input impedance in the first passband of the second filter circuit  36  as seen from the balanced ports P 2 - 2 , P 2 - 3  under the condition where the first terminal  16  of the switch  14  is disconnected from the second terminal  20  and the junctions J 1 , J 2  are opened, has an opposite polarity than, and substantially the same amplitude as, the imaginary component of an input impedance in the first passband of the matching circuit  38  alone as seen from the output terminals  40 - a,    40 - b.  Accordingly, the effect of the second bandpass filter  48  on the impedance matching occurring when the first bandpass filter  46  is being actuated can also be canceled by the matching circuit  38 ; and the impedance matching in the first passband can be maintained 
         [0036]    Therefore, in an embodiment, the phase of a reflection coefficient in the second passband of the first bandpass filter  46  as seen from the balanced ports P 3 - 2 , P 3 - 3  under the condition where the first terminal  16  of the switch  14  is disconnected from the second terminal  18  and the junctions J 1 , J 2  are opened, is compared to the phase of a reflection coefficient in the first passband of the second bandpass filter  48  as seen from the balanced ports P 4 - 2 , P 4 - 3  under the condition where the first terminal  16  of the switch  14  is disconnected from the third terminal  20 , the junctions J 1 , J 2  are opened, and the phase shifters  50 ,  52  are removed. The phase shifters  50 ,  52  are provided to the filter circuit having the filter element of which the phase of the reflection coefficient is leading by 0° to 180°. For example, in the examples shown in  FIGS. 2 to 5 , the phase of the reflection coefficient in the first passband of the second bandpass filter  48  shown in  FIG. 3  (corresponding to the phase of the marker m 2  in  FIG. 3 ) is leading by about 50° the phase of the reflection coefficient in the second passband of the first bandpass filter  46  shown in  FIG. 2  (corresponding to the phase of the marker m 1  in  FIG. 2 ) (that is, the phase of the marker m 2  in  FIG. 3  lies at the position of the marker m 1  in  FIG. 2  rotated counterclockwise by about 50°). The phase shifters  50 ,  52  are connected to the second bandpass filter  48  having a reflection coefficient whose phase is relatively leading. Which of the phases of the bandpass filter  46  and the bandpass filter  48  is leading depends on the specific configurations of these bandpass filters. Accordingly, unlike the examples shown in  FIGS. 2 and 3 , in the case where the phase of the reflection coefficient in the second passband of the bandpass filter  46  is leading the phase of the reflection coefficient in the first passband of the bandpass filter  48 , the phase shifters are provided to the first filter circuit  34  having the bandpass filter  46 . 
         [0037]      FIGS. 2 and 3  show examples wherein both the phase of the reflection coefficient in the second passband of the first bandpass filter  46  (the phase of the marker m 1 ) and the phase of the reflection coefficient in the first passband of the second bandpass filter  48  (the phase of the marker m 2 ) lie in the range from −180° to 0°. The phase of the reflection coefficient of each bandpass filter may be various values. Other examples of the reflection coefficients of the first bandpass filter  46  and the second bandpass filter  48  will now be described with reference to  FIG. 6 .  FIGS. 6   a  to  6   d  schematically show various examples wherein the configurations and passbands of the first bandpass filter  46  and the second bandpass filter  48  are varied. In these examples, the phase of the reflection coefficient in the first passband of the second bandpass filter  48  (denoted by the marker m 2 ) as seen from the balanced ports P 4 - 2 , P 4 - 3  under the condition where the first terminal  16  of the switch  14  is disconnected from the third terminal  20 , the junctions J 1 , J 2  are opened, and the phase shifters  50 ,  52  are removed, is leading by 0° to 180° the phase of the reflection coefficient in the second passband of the first bandpass filter  46  (denoted by the marker m 1 ) as seen from the balanced ports P 3 - 2 , P 3 - 3  under the condition where the first terminal  16  of the switch  14  is disconnected from the second terminal  18  and the junctions J 1 , J 2  are opened As  FIGS. 2 and 3 ,  FIG. 6   a  shows an example wherein the phase of the marker m 1  lies in the range from −180° to −90°, and the phase of the marker m 2  lies in the range from −90° to 0°.  FIG. 6   b  shows an example wherein the phase of the marker m 2  lies in the range from 0° to 90°, and the phase of the marker m 1  lies in the range from −90° to 0°.  FIG. 6   c  shows an example wherein the phase of the marker m 2  lies in the range from 90° to 180°, and the phase of the marker m 1  lies in the range from 0° to 90°.  FIG. 6   d  shows an example wherein the phase of the marker m 2  lies in the range from −180° to −90°, and the phase of the marker m 1  lies in the range from 90° to 180°. In any of the examples shown, the phase of the marker m 2  is leading, by 0° to 180°, the phase of the marker m 1 . This phase difference, therefore, is adjusted by providing phase shifters to the second filter circuit  36  including the second bandpass filter  48  corresponding to the marker m 2 .  FIGS. 6   a  to  6   d  show examples wherein the markers m 1 , m 2  are in different quadrants on the Smith charts. The markers m 1 , m 2  may be in the same quadrant. For example, both the markers m 1 , m 2  may be in the range from −90° to 0°. Thus, in the case where the marker m 1  and the marker m 2  are in the same quadrant, the phase difference between these markers is in the range from 0° to 90°. 
         [0038]      FIG. 7   a  and  FIG. 7   b  are Smith charts showing the frequency characteristics of an input impedance of a front-end module taken when the first bandpass filter  46  is being actuated.  FIG. 8   a  and  FIG. 8   b  are Smith charts showing the frequency characteristics of an input impedance of the front-end module taken when the second bandpass filter  48  is being actuated.  FIG. 7   a  and  FIG. 8   a  show simulations of the front-end module  10  shown in  FIG. 1  freed of the phase shifters  50 ,  52 .  FIG. 7   b  and  FIG. 8   b  show simulations of the front-end module  10  shown in  FIG. 1 . In each figure, the chart on the left shows an input impedance of the respective module as seen from the output terminals  40 - a,    40 -b; and the chart on the right shows an input impedance of the respective module as seem from the antenna terminal  12 . In these figures, the central frequency of the passband of the first bandpass filter  46  is denoted by the marker m 2  or the marker m 4 , and the central frequency of the passband of the second bandpass filter  48  is denoted by the marker m 1  or the marker m 3 .  FIG. 7   a  and  FIG. 8   a  show that, in a front-end module not having the phase shifters  50 ,  52 , the markers m 2 , m 4  are off the reference impedance. In contrast,  FIG. 7   b  and  FIG. 8   b  show that, in a front-end module  10  according to one embodiment of the present invention, the markers m 2 , m 4  are matched with the reference impedance. 
         [0039]    The circuit configuration of the front-end module  10  shown in  FIG. 1  can be varied as necessary. For example, the passbands of the first and second bandpass filters  46 ,  48  described in this specification are examples; and filters having various passbands can be used in place of these filters. Further, the first filter circuit  34  may have another circuit element in addition to the first bandpass filter  46 . For example, phase shifters respectively connected to the balanced ports P 3 - 2 , P 3 - 3  may be provided subsequently to the first bandpass filter  46 . In this case, the matching circuit  38  is configured such that the imaginary component of an input impedance in the second passband of the first filter circuit  34  as seen from the junctions J 1 , J 2  has an opposite polarity than, and substantially the same amplitude as, the imaginary component of an input impedance in the second passband of the matching circuit  38  as seen from the output terminals  40 - a,    40 - b.    
         [0040]    The configurations of the matching circuit  38  explicitly described in this specification are mere examples; the matching circuit  38  can be configured by, for example, combining passive elements such as a capacitor or an inductor in various embodiments. The input impedance of the matching circuit  38  can be adjusted desirably by adjusting the element values of the passive elements constituting the matching circuit  38 . Further, a desired number of bandpass filters can be provided to the front-end module  10  according to the present invention; for example, three or more bandpass filters may be disposed in parallel subsequently to the switch  14 . The front-end module according to the present invention can be installed in various wireless communication devices other than mobile phones. The front-end module according to the present invention can be downsized when it is constructed on a low temperature cofired ceramics (LTCC) multilayered circuit board. Embodiments of the present invention are not limited to those explicitly described above. The embodiments described in this specification are susceptible of various modifications within the purport of the present invention.