Abstract:
An automatic frequency tuning system and method for a transmit power amplifier. The transmit power amplifier has an antenna feed line including a series capacitor and is coupled to an output of an output driver. In one embodiment, the system includes: (1) a shunt capacitor array having a plurality of capacitors selectably couplable to the antenna feed line to apply a programmable shunt capacitance thereto, (2) a peak detector circuit couplable to nodes of the antenna feed line associated with both terminals of the series capacitor and (3) a processor configured to control the peak detector circuit to determine a ratio of voltage levels measured at the nodes at a given power level of the output driver.

Description:
TECHNICAL FIELD OF THE INVENTION 
       [0001]    The invention is directed, in general, to radio frequency (RF) transmitters and, more specifically, to an automatic frequency tuning system and method for a transmit power amplifier. 
       BACKGROUND OF THE INVENTION 
       [0002]    The FM band (i.e., a frequency band between about 76 and about 108 MHz) is used worldwide. FM-band transmitters find broad application in many types of wireless devices, including mobile cellular devices. 
         [0003]    The Q factor, or quality factor, is a dimensionless ratio relating the resonant frequency of a device or circuit to its power dissipation; the higher the Q factor, the greater the efficiency. The Q factor of a loop antenna, such as may be printed on a printed circuit board, can be high. However, its Q factor decreases as frequency decreases. A quarter-wavelength FM-band printed loop antenna would have to be about 75 cm long, which is impractical for mobile cellular devices. The efficiency of a loop antenna that is practical in such devices is relatively low for the FM band. 
         [0004]    While the Q factor is low, it still varies materially over the 76 MHz to 108 MHz range of the FM band. This causes the power of the transmitted signal to vary from one end of the band to the other, which is disadvantageous. Unfortunately, decreasing the antenna&#39;s Q factor to stabilize the power decreases its efficiency and increases its power dissipation. This is disadvantageous in battery-powered devices. 
       SUMMARY OF THE INVENTION 
       [0005]    One aspect of the invention provides an automatic frequency tuning system for a transmit power amplifier. The transmit power amplifier has an antenna feed line including a series capacitor and is coupled to an output of an output driver. In one embodiment, the system includes: (1) a shunt capacitor array having a plurality of capacitors selectably couplable to the antenna feed line to apply a programmable shunt capacitance thereto, (2) a peak detector circuit couplable to nodes of the antenna feed line associated with both terminals of the series capacitor and (3) a processor configured to control the peak detector circuit to determine a ratio of voltage levels measured at the nodes at a given power level of the output driver. 
         [0006]    Another aspect of the invention provides an automatic frequency tuning method for a transmit power amplifier as structured above. In one embodiment, the method includes: (1) setting a synthesizer frequency to a desired FM-band channel frequency, (2) setting the output driver at a power level at which insubstantial leakage occurs in bands other than the FM band, (3) finding a shunt capacitor setting that corresponds to a minimum voltage level at a first node between the output driver and the series capacitor, (4) increasing the power level to a level at which the peak detector circuit is capable of sensing differences between measurement points close to a voltage gain peak, (5) computing ratios of measured voltage levels at the first node and a second node across the series capacitor from the first node for a plurality of shunt capacitor settings and (6) programming the shunt capacitor array with a shunt capacitor setting associated with one of the ratios. 
         [0007]    In another embodiment, the method includes: (1) shorting inputs of a peak detector circuit together, (2) measuring a DC offset of a sense amplifier of the peak detector circuit, (3) setting a synthesizer frequency to a desired FM channel frequency, (4) setting a power level of an FM transmitter to a level at which insubstantial leakage occurs in bands other than the FM band, (5) connecting an input of the peak detector circuit to a first node between an output driver and a series capacitor, (6) adjusting an attenuation ratio of a capacitor divider at an input of the peak detector circuit, (7) sweeping shunt capacitor settings across a range of settings, (8) finding a minimum of voltage measurements, (9) increasing the power level to a level at which the peak detector circuit can sense differences between the measurement points proximate a voltage gain peak, (10) sweeping shunt capacitor settings about the voltage gain peak and (11) computing a voltage level ratio at each of the shunt capacitor settings. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0008]    For a more complete understanding of the invention, reference is now made to the following descriptions taken in conjunction with the accompanying drawings, in which: 
           [0009]      FIG. 1  is a diagram of one embodiment of an automatic frequency tuning system for a transmit power amplifier; 
           [0010]      FIG. 2  is a graph of impedance-matching network voltage gains and peak voltages across a series capacitor of the transmit power amplifier as a shunt capacitor array thereof is swept through each of a plurality of settings; and 
           [0011]      FIG. 3  is a flow diagram of one embodiment of an automatic frequency tuning method for a transmit power amplifier. 
       
    
    
     DETAILED DESCRIPTION 
       [0012]    In the following discussion, “on-chip” denotes components that share the same substrate with the majority of the transmit power amplifier, and “off-chip” denotes components that do not share the same substrate with the majority of the transmit power amplifier and therefore are merely electrically coupled to the remainder of the transmit power amplifier. Internal circuits use on-chip components. 
         [0013]    An internal impedance-matching network associated with a transmit power amplifier may be employed to adjust the Q factor of a loop antenna coupled to the output of the amplifier. To provide proper operation over the FM band, the impedance-matching network can be adjusted as the transmit frequency changes such that the output of the amplifier matches to the antenna impedance at the frequency. More specifically, the impedance-matching network of the transmit power amplifier in a transmitter may be appropriately adjusted to maximize the power of the transmitted signal and maximally attenuate higher harmonics of the transmitted signal. 
         [0014]    While any adjustable impedance-matching network may be employed to tune a loop antenna, one embodiment of a network to be described herein is internal and implemented with capacitors, as on-chip inductors are not possible for FM frequencies due to their low Q factors. In one embodiment, Q factors of on-chip capacitors are higher than the Q factor of the antenna in order to reduce the loss and increase the efficiency of the transmit power amplifier. In one embodiment, metal-insulator-metal capacitors are used to enhance the linearity of the network. Because antenna characteristics may vary from one device to another, one embodiment of the impedance-matching network provides tuning that is wide enough to compensate for device-dependent antenna characteristics as well as provide the tuning required for the FM band, i.e., from 76 MHz to 108 MHz. Antenna impedance may change in response to changes in ambient temperature or electromagnetic field caused by, e.g., humidity, a human body or a magnetic device proximate the antenna. Therefore, one embodiment tunes impedance occasionally, and perhaps periodically, during device normal operation to account for changes in antenna impedance. 
         [0015]    Finally, certain embodiments of the adjustable impedance-matching network function in FM transmitters that operate concurrently with other wireless radios in the same mobile cellular device. Accordingly, leakage of the transmitted FM signal to the frequency bands of these other wireless radios should be within the allowed limits even while the impedance-matching network is being tuned in order not to interrupt any of the other wireless radios&#39; activities. 
         [0016]    Given the above, described herein are various embodiments of a capacitive impedance-matching network that matches the output of a transmit power amplifier to an antenna as the frequency of operation changes. In certain embodiments, the tuning range of the impedance-matching network is wide enough to cover all the internal capacitance variances and antenna impedance variances. 
         [0017]      FIG. 1  is a diagram of one embodiment of an automatic frequency tuning system for a transmit power amplifier.  FIG. 1  shows an output driver  110  coupled to an antenna  140  via an antenna feed line  120  that includes a series capacitor Cm. The output driver  110  typically receives its input signal from a filter (not shown), which in turn receives its input signal from a synthesizer (also not shown). The antenna  140 , which is a printed loop antenna in the illustrated embodiment, is straightforwardly modeled as a capacitance Cp, a resistance Rloss and an inductance L. 
         [0018]    A shunt capacitor array  130  is also coupled to the antenna feed line  120 . The shunt capacitor array  130  contains an array of capacitors that may be selectively coupled in parallel to shunt the antenna feed line  120  to ground (shown but not referenced) with a programmable capacitance. The specific embodiment of the shunt capacitor array  130  of  FIG. 1  shows seven shunt capacitors C 0 , . . . , C 6 , although different numbers of shunt capacitors are employed in alternative embodiments. In the illustrated embodiment, the shunt capacitors are related to one another as powers of two: C 0  is the smallest capacitance, and C 6  has 26 times as much capacitance as C 0 . 
         [0019]    Each of the shunt capacitors C 0 , . . . , C 6 , is selectively closed or opened by way of a corresponding state bit B 0 , . . . , B 6  provided to a corresponding bit input of a first inverter  131   0 , . . . ,  131   6  that, in turn, drives a second inverter  132   0 , . . . ,  132   6 . If a state bit B 0 , . . . , B 6  is zero, a corresponding p-channel metal oxide semiconductor (PMOS) field-effect transistor (FET)  133   0 , . . . ,  133   6  is turned on, and the antenna feed line  120  is weakly coupled to the voltage rail through the corresponding shunt capacitor C 0 , . . . , C 6  and a corresponding resistor  134   0 , . . . ,  134   6 . A corresponding n-channel metal oxide semiconductor (NMOS) FET  135   0 , . . . ,  135   6  is turned off, isolating the antenna feed line  120  from ground. If a state bit B 0 , . . . , B 6  is one, the corresponding NMOS FET  135   0 , . . . ,  135   6  is turned on, and the antenna feed line  120  is strongly coupled to ground through a corresponding shunt capacitor C 0 , . . . , C 6 . The corresponding PMOS FET  133   0 , . . . ,  133   6  is turned off, isolating the antenna feed line  120  from the voltage rail. Thus, a capacitor network can be established that couples the antenna feed line  120  to one or both of the voltage rail and the ground via various of the shunt capacitors C 0 , . . . , C 6 . The state bits B 0 , . . . , B 6  may be thought of as constituting a shunt capacitor setting. 
         [0020]    A peak detector circuit  150  is also coupled to the antenna feed line  120  at a node X and a node Y, which straddle the series capacitor Cm. The peak detector circuit  150  is coupled to the nodes X and Y via capacitors C 0 . Switches SW X1  and SW X2  are controlled to couple the capacitor C 0  that is coupled to the node X either to a noninverting input of a sense amplifier  151  or ground. Switches SW Y1  and SW Y2  are controlled to couple the capacitor C 0  that is coupled to the node Y to the noninverting input of the sense amplifier  151  or ground. Various unreferenced switches are controlled to couple the noninverting input of the sense amplifier  151  to ground, either indirectly through an adjustable input capacitor array C E0 , C E1  or C E2 , or directly in the case of a switch SW ENGND . The noninverting input of the sense amplifier  151  is routed through a bias resistor R bias  and provided as a differential output OUTM to a first differential input of an analog-to-digital converter (ADC)  160 . An inverting input of the sense amplifier  151  is provided as a differential output OUTP to a second differential input of the ADC  160  and further controllably coupled to the voltage rail by a switch SW quickcharge . An NMOS FET  152  is coupled to an inverting output of the sense amplifier  151 , applying current from a current source  153  to the inverting input and OUTP based on the state of the inverting output. An NMOS FET  154  capacitively couples OUTP to ground. A calibration switch SW cal  closes to allow the sense amplifier  151  to be calibrated. 
         [0021]    The ADC  160  provides a digital signal to a digital processing circuit  170 . The digital processing circuit  170 , which may include a processor of any suitable type or capability, is coupled to the shunt capacitor array  130  and the peak detector circuit  150 , namely the various switches therein, to open and close the switches to perform various functions. In a manner to be described below, the digital processing circuit  170  employs the shunt capacitor array  130  and the peak detector array  150  to determine how best to configure the shunt capacitor array to match the output driver  110  to the antenna  140  at a particular FM frequency. 
         [0022]    In the illustrated embodiment, the shunt capacitor array  130  is adjusted as transmit frequency or the antenna impedance changes by automatically adjusting the total shunt capacitor value from the antenna feed line  120  to ground. The series capacitors C 0  behave like an AC coupling capacitor, allowing the sense amplifier  151  to be biased separately. To find the appropriate bit setting B 0 , . . . , B 6  for the shunt capacitor array  130 , the two ends of the series capacitor Cm are connected to the peak detector circuit  150  through the capacitors C 0 . Furthermore, along with the parasitic capacitor of the sense amplifier  151  and the adjustable input capacitor array C E0 , C E1 , C E2 , the two separate series capacitors C 0  act like a capacitive attenuator, which improves the dynamic range of the peak detector circuit and causing it to act like a companding system. Through a feedback loop, the sense amplifier  151  senses the negative peaks of the FM signal provided by the output driver  110 . The time-constant of the peak detector circuit  150  depends on the total impedance on the output provided to OUTP. 
         [0023]    The FM transceiver with which the output driver  110  is associated is a half-duplex system. Thus, the receiver path (not shown) is not active while the output driver  110  is on. Consequently, using the ADC  160  to calibrate the FM transmitter will not interrupt any other activities on the ADC  160  in the illustrated embodiment. In one embodiment, the ADC  160  is used to receive other analog input signals during the transmit operation. However, this embodiment complicates the periodic calibration of the power amplifier during the normal transmit operation. The illustrated embodiment does not use the ADC  160  to receive other analog input signals during the transmit operation. The digital processing circuit  170  processes the output of the ADC  160  digitally to define optimum shunt capacitor settings for the shunt capacitor array  130 . 
         [0024]    As described above, the two ends of the series capacitor Cm of the impedance-matching network, shown as the node X and the node Y in  FIG. 1 , are connected to the input of the peak detector circuit  150  through the capacitors C 0  and the switches SW X1 , SW X2 , SW Y1 , SW Y2 . The values of the capacitors C 0  have been chosen to be close to the finest capacitor value C 0  in the shunt capacitor array  130  to prevent any degradation on the performance of the transmit power amplifier. During the normal operation of the transmitter, the switches SW X2  and SW Y2  are turned on in order to connect the capacitors C 0  to ground and prevent them from floating. Similarly, SW ENGND  is closed to connect the input of the sense amplifier  151  to ground to prevent its intrinsic capacitor from floating as well. Floating capacitors would create nonlinearities and degrade the performance of the adjustment of the shunt capacitor array  130  and ultimately the performance of the FM transmitter. 
         [0025]    During the calibration cycle of the FM transmitter, the DC offset calibration of the sense amplifier  151  is initially determined. To measure the DC offset, its noninverting and inverting inputs are shorted together by enabling the switch SW cal . During this calibration stage, the switches SW X1 , SW Y1  and SW ENGND  are opened. During the calibration cycle of the transmitter, if the node X is connected to the peak detector circuit  150 , the switches SW X1 , SW Y2  are closed, and the switches SW X2 , SW Y1 , SW ENGND  are opened. Similar to the operation explained above, the switch SW Y2  is closed in order to prevent the capacitor C 0  coupled to the node Y from floating. The states of the switches associated with the input capacitor array C E0 , C E1 , C E2  at the input of the sense amplifier  151  are decided according to the required attenuation ratio depending on the output signal swing of the output driver  110 . 
         [0026]    Similarly, during the calibration cycle of the FM transmitter, if the node Y is connected to the peak detector circuit  150 , the switches SW X2 , SW Y1  are closed, and the switches SW X1 , SW Y2 , SW ENGND  are opened. 
         [0027]    The frequency tuning method described herein aims to increase the voltage gain between the nodes X and Y in  FIG. 1 , which is the voltage gain of the impedance-matching network. This maximizes the voltage across the antenna  140  and the linearity of the transmitted output power. 
         [0028]      FIG. 2  is a graph of impedance-matching network voltage gains and peak voltages across a series capacitor of the transmit power amplifier as the shunt capacitor array  130  is swept through each of a plurality of settings. The upper waveform shows the impedance-matching network gain as the shunt capacitor value is swept. The lower waveforms show the peak of the voltages at the nodes X and Y in  FIG. 1 . As  FIG. 2  shows, the peak voltage at the node Y, which provides maximum output power, does not necessarily match with the peak of impedance-matching network voltage gain. However, as mentioned above, the aim is to transmit as much power as possible without creating substantial nonlinearities. Maximizing the voltage gain of the impedance-matching network ensures the maximum output power as well. As an example, as it is seen with the line  210 , the maximum impedance-matching network gain is about 3.8 and if we assume a maximum internal peak voltage of about 550 mV, the output voltage peak is about 2.09 V, which is higher than the maximum voltage peak at the node Y in  FIG. 2 . 
         [0029]    Another complication during the tuning procedure is the power leaking into other wireless radios&#39; frequency bands, which especially occurs if the internal signal swing goes above some threshold voltage, which is close to Vdd/2-Vdsat, where Vdsat is the drain-source saturation voltage of the PMOS FETs in the output driver  110  of  FIG. 1 , the distortion of the transmitted signal increases, which causes an increase in the leakage power. Hence, the transmitted power level during the tuning sequence should be sufficiently low not to cause any degradation in the performance of other wireless radios (e.g., operating in the GSM band). However, if the transmitted signal power is too low, the peak detector circuit  150  may not capture differences between data points close to the minima of the voltage peaks at the node X, where the impedance-matching network voltage gain reaches its maxima. Thus, the transmitted power level during the tuning sequence should be adjusted appropriately. 
         [0030]    With these considerations in mind, described herein are various embodiments of an automatic tuning method that tunes the impedance-matching network automatically for the required transmit frequency. Certain embodiments of the method are performed before the normal operation of the FM transmitter is activated. Other embodiments of the method are performed occasionally and periodically during the normal operation of the FM transmitter in order to adjust to varying operating conditions of the antenna. Still other embodiments of the method ensure that the leakage of the transmitted power to the other wireless radios&#39; frequency operation bands are within the specified limits in order to prevent any interruption on the activities of those wireless radios. 
         [0031]      FIG. 3  is a flow diagram of one embodiment of an automatic frequency tuning method for a transmit power amplifier. The method begins in a start step  305 . In a step  310 , the inputs of the peak detector circuit  150  are shorted together in order to measure the DC offset of the sense amplifier of the peak detector circuit  150 . In a step  315 , the synthesizer frequency is set to a desired FM channel frequency. In one embodiment, the synthesizer is modulated. However, in the illustrated embodiment, the synthesizer is not modulated. 
         [0032]    In a step  320 , the power level of the FM transmitter is set to a low enough level such that there is no substantial leakage in the other wireless radios&#39; bands. In a step  325 , the input of the peak detector circuit  150  is connected to the node X in  FIG. 1 . In a step  330 , the attenuation ratio of the capacitor divider at the input of the peak detector circuit  150  is adjusted according to the transmit power level. 
         [0033]    In a step  335 , the shunt capacitor settings are swept across a range of settings. In the embodiment of  FIG. 1  in which seven state bits, B 0 , . . . , B 6  are used to configure the shunt capacitor array  130 , the shunt capacitor settings are swept from 0 to 127. In one embodiment, the shunt capacitor settings are swept by a step size of eight. However, the step size differs in other embodiments. 
         [0034]    In a step  340 , the minimum of the measurements is found. For purposes of this description, the minimum of the measurements is labeled “Cmincoarse.” Since the relative values between each measurement are being compared, the accuracy of the attenuation ratio is not important. In a step  345 , the power level of the FM transmitter is increased to a high enough level such that the peak detector circuit  150  can sense the differences between the measurement points close to the voltage gain peak. In a step  350 , the shunt capacitor settings are swept about the voltage gain peak. In one embodiment, the shunt capacitor settings are swept from Cmincoarse−8 to Cmincoarse+8 by a step size of one. However, the range and step size of the sweeping differ in other embodiments. 
         [0035]    In a step  350 , at each measurement step, the following steps are repeated. In a step  355 , the value of the attenuation ratio is adjusted. In a step  360 , the input of the peak detector circuit  150  is connected to the node X and the voltage level measured. In a step  365 , the value of the attenuation ratio is adjusted again. In a step  370 , the input of the peak detector circuit  150  is connected to the node Y and the voltage level measured. In a step  375  the ratio between the measurement values determined in the steps  360  and  370  is computed. 
         [0036]    In a step  380 , the shunt capacitor setting corresponding to the maximum ratio is determined, and the shunt capacitor array  130  is set to that shunt capacitor setting by setting the state bits B 0 , . . . , B 6  accordingly. The method ends in an end step  385 . 
         [0037]    Turning back to  FIG. 1  but with continuing reference to  FIG. 3 , the reactance of the antenna  140  typically varies with changing environmental conditions. While the entire tuning range of the shunt capacitor array  130  is necessary to handle all transmission frequencies and process variations on-chip, the magnitude of the reactance change with environmental conditions is often substantially smaller. Thus, for an incremental re-tuning of the FM transmitter, only the steps  350  through  380  of the method of  FIG. 3  need to be re-executed, where the center code (Cmincoarse) is replaced with the previously found shunt capacitor setting. 
         [0038]    Those skilled in the art to which the invention relates will appreciate that other and further additions, deletions, substitutions and modifications may be made to the described embodiments without departing from the scope of the invention.