Abstract:
Method and apparatus for extending a zero voltage switching (ZVS) range during power conversion. In one embodiment, the apparatus comprises a DC/DC converter, operated in a quasi-resonant mode, comprising a transformer; a primary switch, coupled to a primary winding of the transformer, for controlling current flow through the primary winding; and a component coupled to the transformer, wherein the component has a capacitance that varies with voltage across the component, and wherein during a downswing in voltage across the primary switch the component is passively tuned by a change in the voltage across the component that changes the capacitance of the component, and wherein the passive tuning of the component causes a resonant frequency of the DC/DC converter to change, and wherein the change in the resonant frequency causes the downswing to accelerate.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This application is a continuation of co-pending U.S. non-provisional patent application Ser. No. 12/383,497 filed Mar. 25, 2009, which claims benefit of U.S. provisional patent application Ser. No. 61/070,799, entitled “Apparatus for Extending Zero-Voltage Switching Range in a DC to DC Converter”, filed Mar. 26, 2008, both of which are herein incorporated in their entirety by reference. 
     
    
     BACKGROUND OF THE INVENTION 
       [0002]    1. Field of the Invention 
         [0003]    Embodiments of the present invention generally relate to power conversion and, more particularly, to an apparatus for extending the zero-voltage switching (ZVS) range in a DC/DC converter. 
         [0004]    2. Description of the Related Art 
         [0005]    A common topology for DC/DC converters is to operate a flyback converter in a quasi-resonant mode, where the primary switch is activated at the valley of the drain voltage (i.e., a minimum point in the drain-source voltage). The quasi-resonant flyback is a variation of the hard switched flyback, which utilizes the parasitic capacitance of the switch, or even an added capacitance, to absorb leakage inductance energy resulting from a leakage inductance of the DC/DC converter transformer. In addition, by adequately choosing the activation time of the switch, it is possible to have a zero-voltage switching (ZVS) activation characteristic, as well as ZVS deactivation characteristic, in order to improve overall efficiency. 
         [0006]    One issue with such an approach is that a true ZVS transition only occurs in a limited input voltage range and cannot be achieved for all operating conditions. For example, the secondary reflected voltage has to be higher than the input voltage to have a ZVS activation. If such conditions are not met, the energy stored in the capacitance around the primary switch is wasted as the voltage across the primary switch is re-set when the switch turns on, leading to a significant loss of efficiency. 
         [0007]    Therefore, there is a need in the art for the ability to extend the ZVS range in DC/DC converters. 
       SUMMARY OF THE INVENTION 
       [0008]    Embodiments of the present invention generally relate to a method and apparatus for extending a zero voltage switching (ZVS) range during power conversion. In one embodiment, the apparatus comprises a DC/DC converter, operated in a quasi-resonant mode, comprising a transformer; a primary switch, coupled to a primary winding of the transformer, for controlling current flow through the primary winding; and a component coupled to the transformer, wherein the component has a capacitance that varies with voltage, and wherein during a downswing in voltage across the primary switch the component is passively tuned by a change in voltage across the component that changes the capacitance of the component, and wherein the passive tuning of the component causes a resonant frequency of the DC/DC converter to change, and wherein the change in the resonant frequency causes the downswing to accelerate. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0009]    So that the manner in which the above recited features of the present invention can be understood in detail, a more particular description of the invention, briefly summarized above, may be had by reference to embodiments, some of which are illustrated in the appended drawings. It is to be noted, however, that the appended drawings illustrate only a typical embodiment of this invention and are therefore not to be considered limiting of its scope, for the invention may admit to other equally effective embodiments. 
           [0010]      FIG. 1  is a schematic diagram of a DC/DC converter in accordance with one or more embodiments of the present invention; 
           [0011]      FIG. 2  is a graphical diagram of a drain-source voltage V ds  across a primary switch in accordance with one or more embodiments of the present invention; 
           [0012]      FIG. 3  is a schematic diagram of a DC/DC converter in accordance with one or more embodiments of the present invention; and 
           [0013]      FIG. 4  is a flow diagram of a method for increasing a zero voltage switching (ZVS) range in accordance with one or more embodiments of the present invention. 
       
    
    
     DETAILED DESCRIPTION 
       [0014]      FIG. 1  is a schematic diagram of a DC/DC converter  120  in accordance with one or more embodiments of the present invention. In some embodiments, the DC/DC converter  120  may be a flyback converter operated in a quasi-resonant mode; alternative embodiments may comprise different types of DC/DC converters, such as a buck converter, a boost converter, a buck-boost converter, and the like, operated in a quasi-resonant mode. The DC/DC converter  120  may be employed in a stand-alone configuration for DC/DC power conversion, or may be utilized with or as a component of other power conversion devices, such as a DC/AC inverter  124  as shown in  FIG. 1 . The DC/AC inverter  124  additionally comprises a DC/AC conversion module  122 , coupled to the DC/DC converter  120 , for converting an output voltage from the DC/DC converter  120  to an AC output voltage. The DC/AC inverter  124  may be utilized in the conversion of DC power, generated by one or more distributed generators (DGs) such as solar power systems, to AC power. 
         [0015]    The DC/DC converter  120  comprises a capacitor  102  coupled across two input terminals of the DC/DC converter  120  for receiving an input voltage, V in . The capacitor  102  is further coupled across a series combination of a primary winding of a transformer  110  and a semiconductor switch  106  (“primary switch”). The primary switch  106  may comprise one or more switches known in the art, such as metal-oxide-semiconductor field-effect transistors (MOSFETs), bipolar junction transistors (BJTs), emitter switched bipolar transistors (ESBTs), and the like. In some embodiments, a capacitor  108  is coupled across drain and source terminals of the primary switch  106 ; alternatively, the capacitor  108  is not physically coupled across the primary switch  106  but represents parasitic capacitances present at the node, for example capacitances of the semiconductor primary switch  106 , printed circuit board (PCB) capacitances, stray capacitances, and the like. 
         [0016]    A secondary winding of the transformer  110 , having a 1:n turns ratio, is coupled across a series combination of a diode  114  and an output capacitor  116 , with an anode terminal of the diode  114  being coupled to a first terminal of the secondary winding. Two output terminals of the DC/DC converter  120  are coupled across the output capacitor  116  for providing an output voltage, V out . 
         [0017]    In accordance with one or more embodiments of the present invention, a varactor  112  (i.e., a voltage variable capacitor) is coupled across the diode  114 ; alternatively, the varactor  112  may replace the diode  114 . In some embodiments, the varactor  112  has a junction capacitance C var  as follows: 
         [0000]    
       
         
           
             
               
                 
                   
                     C 
                     var 
                   
                   &lt; 
                   
                     
                       C 
                       0 
                     
                     * 
                     
                       
                         ( 
                         
                           1 
                           - 
                           
                             
                               V 
                               var 
                             
                             
                               V 
                               j 
                             
                           
                         
                         ) 
                       
                       M 
                     
                   
                 
               
               
                 
                   ( 
                   1 
                   ) 
                 
               
             
           
         
       
     
         [0018]    where V var  is the varactor voltage and C 0 , V j , and M are coefficients dependent upon the specific varactor employed. The varactor  112  may be comprised of diodes, MOSFETS, BJTs, ceramic capacitors, and the like. 
         [0019]    The DC/DC converter  120  receives the input voltage V in  and converts the input voltage to the output voltage V out . During such conversion, a current I c  flows through the capacitor  102  and a current I p  (“primary current”) is supplied to the primary winding of the transformer  110  in accordance with the timing (i.e., opening and closing) of the primary switch  106 , as driven by a control circuit  104  coupled to a gate terminal of the primary switch  106 . When the primary switch  106  is open, no current flows through the primary winding of the transformer  110  (i.e., I p =0) and the current I c  charges the capacitor  102 . When the primary switch  106  is closed, the capacitor  102  discharges and the primary current I p  increases linearly through the primary winding of the transformer  110 . The primary current I p  additionally flows through a leakage inductance of the transformer  110  that is effectively in series with the primary winding. 
         [0020]    When the primary switch  106  opens, the flow of the primary current I p  through the primary switch  106  ceases and the leakage inductance reverses its voltage, causing a rapid rise of a drain-source voltage V ds  across the primary switch  106  until the threshold voltage of the diode  114  is reached and the diode  114  begins to conduct. As a result of the energy stored in the magnetic field of the transformer  110 , a current I s ˜I P /n (“secondary current”) is induced in the secondary winding and linearly declines to zero. As the secondary current I s  reaches zero, a core reset period begins and the drain-source voltage V ds  begins to ring sinusoidally at a frequency of an LC resonant circuit seen from the primary side comprising the inductance of the primary winding, the capacitance of the capacitor  108 , and the reflected capacitance from the varactor  112 , where the ringing is damped by ohmic losses. 
         [0021]    At the time the secondary current I s  reaches zero, the capacitance of the varactor  112  is large (i.e., approximately C 0 ) due to a varactor voltage V var  close to zero. As the drain-source voltage V ds  begins to swing down, the varactor voltage V var  increases. The rising varactor voltage V var  reduces the varactor capacitance C var , thereby increasing the frequency of the LC resonant circuit during the downward swing of the drain-source voltage V ds  and thus accelerating the downward swing of the drain-source voltage V ds . The accelerated drain-source voltage downswing extends the zero voltage switching (ZVS) range by creating a deeper valley in the drain-source voltage V ds  for the ZVS switching to occur. Thus, the primary switch  106  can be activated at a V ds  closer to zero than that which would be possible without the effect of the varactor  112 . In some embodiments, the ZVS range may experience at least a 30% increase. 
         [0022]      FIG. 2  is a graphical diagram of a drain-source voltage V ds  across a primary switch  106  in accordance with one or more embodiments of the present invention. The primary switch  106  operates within the DC/DC converter  120  as previously described with respect to  FIG. 1 . Prior to T 0 , the primary switch  106  is closed and current flows through the primary switch  106 . At time T 0 , the primary switch  106  opens (i.e., turns off), thereby terminating the flow of current through the primary switch  106 . Additionally, the leakage inductance of the transformer  110  reverses its voltage, causing a rapid rise of the drain-source voltage V ds . Once the threshold voltage of the diode  114  is reached, the diode  114  begins to conduct and a secondary current I s ˜I P /n is induced in the secondary winding and linearly declines to zero. 
         [0023]    At time T 1 , the secondary current I s  reaches zero and the varactor voltage V var  is close to zero, resulting in a large capacitance of the varactor  112  (i.e., approximately C 0 ). A core reset period begins, and the drain-source voltage V ds  begins to ring at the frequency of the LC resonant circuit. 
         [0024]    From time T 1  to T 2 , as the drain-source voltage V ds  begins to decline, the varactor voltage V var  rises and reduces the varactor capacitance C var , thereby increasing the resonant frequency of the LC resonant circuit during the downward swing of the drain-source voltage V ds . From time T 2  to T 3 , the increased resonant frequency accelerates the downward swing of V ds , resulting in a V ds  downswing  202  that is more rapid than a V ds  downswing  204  that would occur when the resonant frequency of the LC circuit remains unchanged (i.e., in the absence of the varactor  112 ). 
         [0025]    The accelerated V ds  downswing  202  results in a lower valley in the drain-source voltage V ds  at time T 3  than a valley which would occur in the absence of the varactor  112 , thus creating an extended ZVS range  206 . The extended ZVS range  206  allows the primary switch  106  to be activated at a lower drain-source voltage V ds  (i.e., V 1 ) than that which would be possible without the effect of the varactor  112  (i.e., V 2 ), resulting in an energy savings of ½C*(V 2 −V 1 ) 2 , where C is the capacitance of the capacitor  108 . 
         [0026]      FIG. 3  is a schematic diagram of a DC/DC converter  120  in accordance with one or more embodiments of the present invention. In some embodiments, the DC/DC converter  120  may be a flyback converter operated in a quasi-resonant mode; alternatively, the DC/DC converter  120  may be a buck converter, a boost converter, a buck-boost converter, or similar type of DC/DC converter. The DC/DC converter  120  may be employed in a stand-alone configuration for DC/DC power conversion, or may be utilized with or as a component of other power conversion devices, such as the DC/AC inverter  124  as shown in  FIG. 3 . Additionally, as previously described, the DC/AC inverter  124  comprises a DC/AC conversion module  122 , coupled to the DC/DC converter  120 , for converting an output voltage from the DC/DC converter  120  to an AC output voltage. The DC/AC inverter  124  may be utilized in the conversion of DC power, generated by one or more distributed generators (DGs) such as solar power systems, to AC power. 
         [0027]    The DC/DC converter  120  comprises a capacitor  302  coupled across two input terminals of the DC/DC converter  120  for receiving an input voltage, V in . The capacitor  302  is further coupled across a series combination of a primary winding of a transformer  310  and a semiconductor switch  306  (“primary switch”). The primary switch  306  may comprise one or more switches known in the art, such as metal-oxide-semiconductor field-effect transistors (MOSFETs), bipolar junction transistors (BJTs), emitter switched bipolar transistors (ESBTs), and the like. A voltage clamp circuit  308 , comprising a diode  318 , a varactor  320 , a capacitor  322 , and a resistor  324 , is coupled across the primary switch  306  for controlling a spike in the drain-source voltage created by leakage inductance energy from the transformer  310 , as further described below. Additionally, a capacitor  312  is shown coupled across the primary switch  306  to represent parasitic capacitances present at the node, such as capacitances of the semiconductor primary switch  306 , PCB capacitances, stray capacitances, and the like. 
         [0028]    An anode terminal of the diode  318  and a first terminal of the varactor  320  are coupled to a drain terminal of the primary switch  306 ; a cathode terminal of the diode  318  and a second terminal of the varactor  320  are coupled to a first terminal of the capacitor  322  and a first terminal of the resistor  324 . A second terminal of the capacitor  322  and a second terminal of the resistor  324  are coupled to a source terminal of the primary switch  306 . In some embodiments, the varactor  320  has a junction capacitance C var  as follows: 
         [0000]    
       
         
           
             
               
                 
                   
                     C 
                     var 
                   
                   &lt; 
                   
                     
                       C 
                       0 
                     
                     * 
                     
                       
                         ( 
                         
                           1 
                           - 
                           
                             
                               V 
                               var 
                             
                             
                               V 
                               j 
                             
                           
                         
                         ) 
                       
                       M 
                     
                   
                 
               
               
                 
                   ( 
                   2 
                   ) 
                 
               
             
           
         
       
     
         [0029]    where V var  is the varactor voltage and C 0 , V j  and M are coefficients dependent upon the specific varactor employed. The varactor  320  may be comprised of diodes, MOSFETS, BJTs, ceramic capacitors, and the like. In one or more alternative embodiments, the varactor  320  may replace the diode  318 . 
         [0030]    A secondary winding of the transformer  310 , having a 1:n turns ratio, is coupled across a series combination of a diode  314  and an output capacitor  316 , with an anode terminal of the diode  314  being coupled to a first terminal of the secondary winding; in some embodiments, the transformer ratio may be below one (i.e., a step-down transformer). Two output terminals of the DC/DC converter  120  are coupled across the output capacitor  316  for providing an output voltage, V out . 
         [0031]    Analogous to the operation previously described, the DC/DC converter  120  receives the input voltage V in  and converts the input voltage to the output voltage V out . During such conversion, a current I c  flows through the capacitor  302  and a primary current I p  is supplied to the primary winding of the transformer  310  in accordance with the timing (i.e., opening and closing) of the primary switch  306 , as driven by a control circuit  304  coupled to a gate terminal of the primary switch  306 . When the primary switch  306  is open, no current flows through the primary winding of the transformer  310  (i.e., I p =0) and the current I c  charges the capacitor  302 . When the primary switch  306  is closed, the capacitor  302  discharges and the primary current I p  increases linearly through the primary winding of the transformer  310 . The primary current I p  additionally flows through a leakage inductance of the transformer  310  that is effectively in series with the primary winding. 
         [0032]    When the primary switch  306  opens, the flow of the primary current I p  through the primary switch  306  ceases and the leakage inductance reverses its voltage, causing a rapid rise of the drain-source voltage V ds  that results in a spike well over the reflected voltage of V out /n. The resistor  324 , capacitor  322 , and diode  318  act as an RCD (resistor/capacitor/diode) clamp to limit such a spike and prevent damage to the primary switch  306 . 
         [0033]    As the drain-source voltage V ds  increases following the opening of the primary switch  306 , the voltage across the diode  314  increases until the threshold voltage is reached and the diode  314  begins to conduct. As a result of the energy stored in the magnetic field of the transformer  310 , a secondary current I s ˜I P /n is induced in the secondary winding and linearly declines to zero. Analogous to the operation previously described with respect to  FIG. 1 , when the secondary current I s  reaches zero the drain-source voltage V ds  begins ringing sinusoidally due to an LC resonant circuit seen on the primary side comprising a capacitive component from the varactor  320 , where the ringing is damped by ohmic losses. As the drain-source voltage V ds  falls, the varactor voltage V var  increases and reduces the varactor capacitance C var . The decreasing varactor capacitance C var  increases the frequency of the LC resonant circuit during the downward swing of the drain-source voltage V ds , resulting in an accelerated downswing of the drain-source voltage V ds . Such an accelerated downswing extends the ZVS range by creating a deeper valley for the ZVS switching to occur. In some embodiments, the ZVS range may experience at least a 30% increase. 
         [0034]      FIG. 4  is a flow diagram of a method  400  for extending a zero voltage switching (ZVS) range in accordance with one or more embodiments of the present invention. The method  400  begins at step  402  and proceeds to step  404 . At step  404 , a DC/DC converter is operated in a quasi-resonant mode. The DC/DC converter comprises a transformer having a 1:n turns ratio and may be a flyback converter, a buck converter, a boost converter, a buck-boost converter, or similar type of DC/DC converter. In some embodiments, the DC/DC converter may be utilized in a stand-alone configuration for DC/DC power conversion; alternatively, the DC/DC converter may be utilized with or as a component of other power conversion devices, such as a DC/AC inverter  124 . Such a DC/AC inverter may be utilized in the conversion of DC power, generated by one or more distributed generators (DGs) such as solar power systems, to AC power. 
         [0035]    At step  406 , a switch (“primary switch”) of the DC/DC converter, coupled in series with a primary winding of the transformer, is activated for generating a current (“primary current”) through the primary winding, and the primary current linearly increases. At step  408 , the primary switch is deactivated and the primary current ceases. Due to a leakage inductance of the primary winding, a drain-source voltage across the primary switch rapidly increases until a diode coupled to the transformer secondary winding is activated and a current (“secondary current”) is induced in the secondary winding. In some embodiments, a spike in the drain-source voltage during such a rapid increase is limited by a voltage clamp circuit coupled to the primary winding. 
         [0036]    The secondary current linearly declines to zero. Once the secondary current reaches zero, the drain-source voltage begins ringing sinusoidally due to an LC resonant circuit of the DC/DC converter, where the ringing is damped by ohmic losses. The method  400  proceeds to step  410 . 
         [0037]    At step  410 , the frequency of the LC resonant circuit is increased during the downward swing of the ringing drain-source voltage, for example by decreasing a capacitance of the LC resonant circuit during this time. In some embodiments, a varactor having a junction capacitance that decreases as the corresponding varactor voltage increases may be utilized to provide a capacitive component of the LC resonant circuit, where the varactor voltage is increased as the drain-source voltage decreases. Such a varactor may be coupled to the secondary winding of the transformer; alternatively, the varactor may be part of the voltage clamp circuit coupled to the primary winding. The increased resonant frequency accelerates the downward swing of the drain-source voltage, creating a deeper valley (i.e., an extended ZVS range) for switching to occur. 
         [0038]    At step  412 , the primary switch is activated at a valley of the drain-source voltage, and a primary current flows through the primary winding as previously described. In some embodiments, the primary switch may be activated at the first valley of the ringing drain-source voltage; alternatively, the primary switch may be activated at a subsequent valley. The method  400  proceeds to step  414 , where a decision is made whether to continue operation of the DC/DC converter. If the result of such decision is yes, the method  400  returns to step  408 ; if the result of such decision is no, the method  400  proceeds to step  416  where it ends. 
         [0039]    While the foregoing is directed to embodiments of the present invention, other and further embodiments of the invention may be devised without departing from the basic scope thereof, and the scope thereof is determined by the claims that follow.