Abstract:
The invention pertains to a switchmode power inverter, and particularly to a half-bridge inverter for asymmetrical load. More particularly, the invention pertains to high frequency electronic ballast for gas discharge devices, especially for high intensity discharge lamps, completed by an internal high voltage ignition circuit and transient current control protecting against the asymmetrical feature of high intensity discharge lamps at startup.

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
   Not applicable 
   BACKGROUND OF THE INVENTION 
   The present invention relates to high frequency switchmode half-bridge inverters for asymmetrical loads and specifically to high frequency electronic ballasts for gas discharge devices. More specifically, the present invention relates to high frequency electronic ballast for high intensity discharge (HID) lamps. 
   The prior art is replete with many known half-bridge inverters providing high frequency ballast for gas discharge lamps, especially for HID lamps. For instance, high efficient electronic ballasts based on half-bridge inverter configuration which can be used with HID (HPS) lamps are U.S. Pat. No. 5,313,143 entitled “Master-slave half-bridge DC-to-AC switchmode power inverter” (See also a paper entitled “Master-Slave Half-Bridge Inverter” presented at APEC&#39;93); U.S. Pat. No. 5,229,927, entitled “Self-symmetrizing and self-oscillating half-bridge power inverter”, and U.S. Pat. No. 6,329,761, entitled “Frequency controlled half-bridge inverter for variable loads” from the same inventor of the present invention. Further applications of half-bridge inverters are U.S. Pat. No. 5,253,157, entitled “Half-bridge inverter with capacitive voltage equalizer” from Severinsky; U.S. Pat. No. 6,242,867, entitled “Circuit for synchronizing the ignition of electronic ballast discharge lamps” from Pogadaev, and U.S. Pat. No. 5,932,976, entitled “Discharge lamp driving” from Maheshwari, especially for ignition methods. Some of the cited inventions provide solution for equalizing the voltages of the voltage divider capacitors of a half-bridge inverter, but none of these inventions solves a specific problem related to the startup process of a HID lamp, especially for metal halide lamps, described in the following part. 
   An important application of the switchmode power inverters is supplying gas discharge devices, especially high intensity discharge (HID) lamps in the range of 35W to 400W. In this case, the load impedance of the inverter is a HID lamp connected in series with an inductor. At high frequency powering of a HID lamp, the interaction between the ballast and the lamp is more sensitive than that of a conventional low frequency (50/60 Hz) ballast. During the startup process, including the transition from glow to arc discharge, HID lamps may have asymmetrical impedance resulting unequal voltages of the voltage divider capacitors of the half-bridge inverter (See  FIG. 8 ). 
   For instance, let V 2 &gt;V 3 , where V 2 +V 3 =V 1 =constant. At the end of the startup process, the lamp goes into arc discharge state having symmetrical impedance. Therefore, if the ON-times of the main switches are equal, a transient process starts, and after a certain time interval, the equilibrium V 2 =V 3  is achieved. During this transient process high current peak occurs (V 2 &gt;V 3 ) which can damage the main switches of the inverter. Furthermore, the inductor may be also saturated causing an extra current peak exceeding the maximum allowable current peak values of the main switches. 
   The present invention provides a protection, namely a dynamic solution for the limitation of high current peaks during the startup process. 
   Furthermore, the present invention introduces a different, and more effective ignition solution than the ignition solution of U.S. Pat. No. 6,329,761, providing essentially less stress for the main switches of a half-bridge inverter. 
   Also, the present invention provides a special MOSFET driver solution for the main switches of a half-bridge inverter, wherein the main switches can be simultaneously switched off applying an more effective solution comparing to the MOSFET drivers applied in the mentioned patent applications, for instance in U.S. Pat. No. 6,329,761, wherein the main switches can be simultaneously switched off slowly, causing high dissipation in the switches at inductive load, therefore, they are incapable for fast and repeating current protection caused by an asymmetrical load during the startup process. 
   Furthermore, the theoretical background for the practical lamp power control, where the lamp is connected in series with an inductor and supplied by a square wave inverter (push-pull, half-bridge, or full bridge) resulting a special ballast curve can be found in a paper of J. Melis, entitled “Ballast Curves for HPS Lamps Operating on High Frequency” (IAS&#39; 92). 
   BRIEF SUMMARY OF THE INVENTION 
   It is an object of the present invention to provide a high efficient switchmode half-bridge square wave inverter which has protection against the effect of asymmetrical loads. 
   A second object of the present invention to provide a reliable electronic ballast for gas discharge devices acting as temporary asymmetrical loads at startup. 
   A further object of the present invention to provide an effective driver solution capable to switch off simultaneously the main switches of a half-bridge inverter controlled by square wave voltage signal having three states (+12V, 0V, −12V). 
   Another object of the present invention to provide a simple power control of the load, especially HID lamps, where the lamp power remains constant during the aging of the lamp which means continuously increasing lamp voltage, or equivalently, increasing ohmic impedance at high frequency operation. 
   Further object of the present invention to provide a fast, direct limitation of the load current caused by the asymmetrical impedance of an ignited high intensity gas discharge (HID) lamp during the startup process. 
   Another object of the present invention to provide a high voltage ignition circuit for a reliable ignition of HID lamps, especially an almost instant reigniting of warmed up lamps in a wide temperature range. 
   Further object of the present invention to provide dimming capability for the lamp providing significant energy saving under certain conditions when the full power (full light) of the lamp is not required in certain times. 
   These and other objects, features and advantages of the present invention will be more readily apparent from the following detailed description, wherein reference is made to the drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  illustrates a schematic diagram of the preferred half-bridge inverter for HID lamps connected to a DC power supply (power factor pre-regulator) including an Ignitor, Current Transformer, two identical MOSFET Drivers, Logic Supply and a Control Unit; 
       FIG. 2  shows the ignition signals and related control waveforms; 
       FIG. 3A  shows the preferred schematic diagram of the preferred MOSFET Drivers providing effective drivers and fast simultaneous switching off solution for the main switches, which are generally accomplished by MOSFETs in higher frequency range; 
       FIG. 3B  shows the control signals of preferred MOSFET Driver; 
       FIG. 4  shows the preferred schematic diagram of the Control Unit; 
       FIG. 5  shows the waveforms of an unsuccessful startup, namely in the cases of no load condition or failed lamp ignition; 
       FIG. 6  shows the waveforms of a normal, successful startup process; 
       FIG. 7  shows the main voltage waveforms of the Control Unit; 
       FIG. 8  shows the waveform of the transient asymmetrical operation during the startup process of a HID lamp; 
       FIG. 9  shows the ballast curve including the warming up and constant power ranges provided by the present invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1  shows a schematic diagram of the preferred half-bridge inverter configuration as an electronic ballast for HID lamps, where the Half-Bridge Inverter, connected to a DC power supply through the capacitor C 1 , is illustrated in detail. In many cases, the DC power supply is implemented by a Power Factor Pre-regulator providing high power factor and stabilized DC voltage source (V 1 ) for the Half-Bridge Inverter as it is also shown in  FIG. 1 . The Input Unit is connected to an AC Power Supply (50/60 Hz, 120V–240V). A Logic Supply provides stabilized 12V for the Control Unit (connecting points  1  and  2 ) which is also shown in  FIG. 1 . The Control Unit will be described in details. 
   The Half-Bridge Inverter includes the basic components of a half-bridge inverter: two electronically controlled switches (MOSFETs T 1  and T 2 ), two voltage divider capacitors C 2  and C 3  and a load impedance (a HID lamp connected in series with an inductor M 3 ). 
   The Half-Bridge Inverter also includes the preferred embodiment of a high voltage ignition apparatus in which winding N 1  of the inductor M 3  is connected in series with the capacitor C 5  and MOSFET T 3 . When T 3  is on, a high frequency damped sinusoidal voltage occurs across the winding N 1 . This voltage is transformed up by winding N 2  to an approximately 2000V providing sufficient ignition voltage (V L ) for a HID lamp shown in  FIG. 2 , also achieving almost instant reigniting of warmed up lamps. The ON/OFF-times of transistor T 3  is controlled by the Control Unit (connecting point  4 ), where V 9  (see  FIG. 4 ) is the gate voltage of T 3 . The capacitor C 5  is periodically charged by resistor R 5  since the OFF-time of T 3  is essentially longer then its ON-time. The rectifier D 1  limits the drain voltage of MOSFET T 3  approximately to the DC supply voltage V 1 . 
   The Half-Bridge Inverter further includes the preferred embodiment of two identical MOSFET drivers MD- 1  and MD- 2  utilized by the present invention. The MOSFET Driver MD- 1  is shown in  FIG. 3A , including a low power MOSFET T 4 , a low power bipolar transistors T 5 , rectifiers D 9 , D 10 , D 11  and D 12  connected in a bridge configuration, a capacitor C 9 , resistors R 18 , R 19 , R 20  and R 21 , and a diode D 12 . A square wave AC control signals V 6  (see  FIG. 3B ) is provided by the secondary winding N 4  (connecting points A 1  and A 2 ) of the low power signal transformer M 2  shown in  FIG. 1 . Similarly, the secondary winding N 5  (connecting points A 3  and A 4 ) is connected to the MOSFET Driver MD- 2 . The primary winding N 3  of the signal transformer M 2  is connected to the connecting points  8  and  9  of the Control Unit. During the positive half-period, with respect to the point sign of the secondary winding N 4 , a positive voltage is connected across the resistor R 19  and rectifier D 10  to the gate of the N-channel power MOSFET T 1  providing ON-state, while the MOSFET T 4  is in OFF-state. During the negative half-period, a positive voltage is connected across resistor R 18  and rectifier D 9  to the gate of T 4  providing ON-state. Therefore, the gate of T 1  is short circuited to the source of T 1  by MOSFET T 4  providing an excellent current sink capability thus a very short switching off time for MOSFET T 1 . In both half periods the capacitor C 9  is charged through the resistor R 20  and diode D 12  nearly to the amplitude of the square wave voltage V 6 . Furthermore, the bipolar transistor T 5  is connected to the gate of MOSFET T 4  in such a way that when the output voltage of the control transformer is zero, the MOSFETs T 4  will be ON for an appropriate time, therefore power MOSFET T 1  will be OFF as it is illustrated in  FIG. 3B  where V 17  is the gate voltage of T 1  and V 18  is the gate voltage of T 4 . This low power loss MOSFET driver was specifically designed for inductive loads as it is in our case. Evidently, the same description can be applied for the upper MOSFET driver MD- 2 . Therefore, very low power loss can be achieved with respect to the switching transistors T 1  and T 2 , resulting high efficiency for the half-bridge inverter. The main signals for the preferred MOSFET driver are shown in  FIG. 3B , illustrating the simultaneous OFF-states of the main switching power transistor T 1  and T 2  if the square wave control signals V 6 =V 7 =0 in current limiting mode. 
   The Half-Bridge Inverter also includes a Current Transformer including an actual transformer M 1 , where the primary winding N 6  is connected in series with the load and the secondary winding N 7  connected to the AC input of a bridge rectifier implemented by the fast rectifiers D 2 , D 3 , D 5 , and D 5 . The DC output of the bridge rectifier is connected to resistor R 6  providing low voltage signal V 5  nearly proportional to the load current. The output points of the Current Transformer A 5 , and A 6  are connected to the connecting points  5  and  6  of the Control Unit as it is shown in  FIG. 1 . Therefore, the transient operation at asymmetrical loads can be controlled by the Control Unit as it will be described later. 
   The Power Unit further includes a power resistor R 1  in which the current I 1  has an unidirectional high frequency waveform determined by the DC Power Supply, which is generally a boost converter. The voltage across the resistor is filtered by C 4  and R 3  connected in series with R 2  (RC filter), therefore the voltage V 4  across C 4  is nearly DC and proportional to the average load current. This voltage—assuming nearly constant DC supply voltage V 1  for the half-bridge inverter, is also proportional to the input power of the half-bridge inverter (ballast) which is nearly equal to the lamp power. Since the input voltage (V 1 ) of the actual ballast unit (Half-Bridge Inverter) is nearly constant, the control of the lamp power can be easily implemented by frequency control in a relatively narrow range. If the Dimming Switch S 1  (connected in series with resistor R 4 ) is ON, the voltage V 4  significantly increases. In this case the frequency of the inverter will have a predetermined maximum value providing approximately half power for the lamp. 
     FIG. 4  shows a detailed schematic diagram of the Control Unit providing appropriate control signals for the Half-Bridge Inverter, namely driver signals for MOSFETs T 1 , T 2 , and T 3 . Functionally, the Control Unit has three basic parts: a Timer, a Current Limiter, and a Frequency Controller connected to a voltage controlled oscillator (VCO) IC 4 . 
   A) Timer. The Timer unit is controlled by voltage comparator IC 1 , where the inverting input is connected through the connecting point  3  of the Control Unit to the common point of the voltage divider resistors R 7  and R 8  (A 7 ) shown in  FIG. 1 . Therefore, the voltage on the inverting input of IC 1  (V 1   p ) is proportional to the voltage V 1  (see  FIG. 1 ). 
   The startup process of the Control Unit ( FIG. 4 ), therefore the whole circuit, is illustrated in  FIG. 5  and  FIG. 6 , where V 1   s &lt;V 1   (nom) &lt;V 1   r &lt;V 1   (max) . 
   1. V 1 ≧V 1   s . The Logic Supply provides stabilized 12V for the Control Unit, and V 8 =12V since V 1   p&lt; 6   V, and V 10 =12V. 
   2. V 1 ≧V 1   r . The voltage V 8 =0 since V 1   p&gt; 6   V, and the TIMER starts providing periodical ON/OFF signals (V 9 ) for T 3 . 
   3a. Unsuccessful ignition or no load condition (see  FIG. 5 ). After a predetermined time, for instance 120 s, the voltage V 10  goes to zero and stops the periodical ON/OFF signal V 9  for T 3  (V 9 =0), and also switches off the main switches T 1  and T 2  (V 15 =V 17 =0). 
   3b. Successful ignition (see  FIG. 6 ). Assuming that the first ignition signal ignited the lamp, the supply voltage V 1  drops bellow V 1   r  (the DC Power Supply is loaded, and V 1 =V 1   (nom)  which is equal to its regulated value), therefore V 8 =12V and it resets the TIMER (V 9 =0, and V 10  remains high). The circuit solution for the Timer may be based on a CMOS ripple counter and a simple square wave oscillator. 
   B) Current Limiter. The output signal V 5  of Current Transformer is connected to the connecting points  5  and  6  of the Control Unit as it is shown in in  FIG. 4 . Therefore, the voltage of the inverting input of the voltage comparator V 12 =V 11 +V 5  as it is shown in  FIG. 7 . The voltage V 11  is provided by the common point of the voltage divider resistors R 9  and R 10 . Three resistors, R 11 , R 12  and R 13  are connected in series, where the common point of resistors R 11  and R 12  is connected to the non-inverting input of IC 2  and the common point of resistors R 12  and R 23  is connected to the output of IC 2  through a diode D 6 , resulting a resettable bistable multivibrator. The reset is provided in every half-period by differentiating the output signals V 14  and V 15  of the IC 4  (VCO). These differentiators include capacitors C 8 , C 7 , and resistors R 16 , R 17 , respectively. The resulting waveforms are added and rectified by diodes D 7 , D 8  and connected to the non-inverting input of IC 2 . The waveform V 13  is shown in  FIG. 7 . Under normal condition when the output current I L  remains in a predetermined range, the circuit has no effect. If the output current reaches the maximum allowable value, the output voltage of IC 2  goes to zero (V 10 =0) as it is shown in  FIG. 7 , and forces the driver signals V 15  and V 17  going to zero since it is connected to the each input of the dual input AND gates (IC 5 / 1 , IC 5 / 2 , IC 5 / 3  and IC 5 / 4 ). The other inputs of the dual input AND gates are connected to the outputs of IC 4  (V 14  and V 15 ). When V 10 =0, than V 15 =V 17 =0 and the main switches T 1  and T 2  of the inverter are simultaneously switched off as it was described previously. All main signals of the Control Unit are summarized in  FIG. 7 . Furthermore,  FIG. 8  also illustrates the operation of the Current Limiter at asymmetrical operation, when V 2 &gt;V 3  (V 2 +V 3 =V 1 , and remains constant) and the output (lamp) current I L  reaches a predetermined maximum value I L(max) . The normal (symmetrical) operation is also shown in  FIG. 8 , where V 2 =V 3 , and the output current I L  is symmetrical. 
   C) Frequency Controller. This unit is based on the operational amplifier IC 3 , where the non-inverting input (connecting point  7 ) is connected to voltage V 4  of capacitor C 4  shown in  FIG. 1 . The output of IC 3  controls the frequency of the voltage controlled oscillator IC 4 . The outputs V 14  and V 16  are symmetrical square wave signals in opposite phase and are connected to the dual input AND gates of IC 5 . The gates IC 5 / 1 , IC 5 / 2  and IC 5 / 3 , IC 5 / 4  are connected parallel for increased current sink and source capability. The outputs (V 15  and V 17 ) of IC 5  are the connecting points  8  and  9  of the Control Unit and are connected to the primary winding (N 3 ) of transformer M 2  (see  FIG. 1 ) resulting a full-bridge configuration. As it was mentioned previously, the voltage V 4  is nearly proportional to the lamp power. Therefore, the lamp power can be controlled by the frequency of a VCO (IC 4 ) in a certain range. The maximum and the minimum frequency is determined by the resistors R 16  and R 17 , respectively. In summary, the frequency of the voltage controlled oscillator IC 4  is controlled by the operational amplifier IC 3  in such a way that the lamp power remains the same in a predetermined lamp voltage range (80V–160V for HPS lamps and 120V–150V for MH lamps). 
     FIG. 9  shows the ballast curve (lamp power P L  vs. lamp voltage V L ). It includes the warm up range (V L &lt;V L(min) ), where the lamp current frequency is minimum (f min ), and the constant power range (V L(min) ≦V L ≦V L(max) ), where the frequency is controlled (f min ≦f≦f min +Δf) providing nearly constant lamp power. It also shows the dimmed operation, where the frequency has its maximum value (f max ) and remains the same (uncontrolled operation). The dimmed operation is achieved by closing the Dimming Switch S 1  shown in  FIG. 1 . Therefore, the voltage V 4  increases significantly causing the operational amplifier IC 3  (see  FIG. 4 ) out of its control range, and the frequency of the VCO (IC 4 ) will have its maximum value determined by the resistor R 14  and R 15 . At dimmed operation the output power (lamp power) is not controlled, but it remains in an acceptable practical range, determined by the ballast curve at maximum frequency. The dimmed operation, where the lamp power is approximately the half (40%–50%) of its nominal value provides significant energy saving if the full lamp power is not required in certain times. 
   Thus, while preferred embodiments of the present invention have been shown and described in details, it is to be understood that such adaptation and modifications as may occur to those skilled in the art may be employed departing from the spirit and scope of the invention, as set forth in the claims.