Abstract:
A multi-rate tracking circuit with an input for a signal, an output arranged to indicate a current proposed level related to the signal, and voting logic connected to the input, arranged to indicate of a direction of change in the current proposed level. A first counter is connected to the voting logic, and arranged to vary the current proposed level based on the indications received from the voting logic. A second counter is arranged to vary a value based on the indications from the voting logic. The variation of the current proposed level by the first counter is dependent on the value varied by the second counter.

Description:
[0001]    This application claims priority under 35 U.S.C. 119(a) to GB Provisional Application No. 0702593.5 filed Feb. 9, 2007. 
         [0002]    This application claims priority under 35 U.S.C. 119(e)(1) to U.S. Provisional Application No. 61/016,876 (TI-63539P) filed Dec. 27, 2007. 
     
    
     BACKGROUND OF THE INVENTION 
       [0003]    1. Field of the Invention 
         [0004]    This invention relates to a multi-rate tracking circuit. 
         [0005]    A receiver circuit will often be designed to take an input that moves across a certain range, for example between −L 2  and L 2  as shown in  FIG. 6 . However, when a data signal is being transmitted to a receiver circuit, the characteristics of the line over which the data is being transmitted will often act to compress the waveform of the digital signal, resulting in a signal that only moves across a smaller range, as shown by signal B in  FIG. 6 , which only moves across the range −L 1  to L 1 . A receiver circuit taking as input such a signal may be prone to errors, or may not work at all. A known solution to this problem is to amplify the input signal so that it moves across the whole desired range. An example of such an amplified input signal is shown by signal A in  FIG. 6 . 
         [0006]    2. Description of Related Art 
         [0007]    In order to amplify the signal by the correct amount, the receiver will incorporate “voting logic”, which considers a number of characteristics of the signal and decides whether the amplification level needs to be increased or decreased.  FIG. 7  shows an example a circuit trying to find an ideal amplification level L*. At time t 0  the amplification begins at a base level L 0 . Between time t 0  and t 1  the amplification level is below the ideal level L*, and so the voting logic will indicate that the amplification needs to be increased. At time t 1  the amplification level reaches the ideal level L*. However, the receiver is not aware that it has hit the correct level, and thus overshoots. However, once it has overshot the voting logic will indicate instead that the amplification level is above the ideal level L*. The amplification level is then reduced until it falls below the ideal level L*, at which point the voting logic indicates again that it is below the ideal level L*. In this way the amplification level will oscillate close to the ideal amplification level L*. 
         [0008]    Receiver circuits will often incorporate circuitry to compensate for other effects the characteristics of line over which data is transmitted have on the signal. For example, a receiver circuit that takes a digital signal may adjust the amplitude of the bits of the signal to allow for inter-symbol interference. Inter-symbol interference is distortion of a symbol caused by symbols transmitted before and after it. An example of inter-symbol interference is shown in  FIG. 8 , which shows a signal D that is sent down a line, and the resulting signal T. As can be seen, the line acts to smooth the corners of the signal D when it changes between the high and low state, and a sequence of three high states in the signal D results in a gently rising waveform in the resulting signal T. A common method of allowing for inter-symbol interference is to adjust the amplitudes of the signal T to recover the signal D. 
         [0009]    If more than one type of compensation is performed on a signal at the same time, it is necessary to ensure that they do not interfere with each other. A known solution is to have the compensation for the compression of the input signal occur over a much longer time frame than the compensation for inter-symbol interference. For example, while the inter-symbol interference compensation operates on each bit of the input signal in turn, the compression compensation might operate on around a million bits of the input signal. This means that the inter-symbol interference compensation will able to operate without incorrectly trying to compensate for amplitude changes due to the compression compensation. 
         [0010]    However, there are two potential problems with having the compression compensation operate over such a long time frame. The voting logic only indicates whether the amplification level is above or below the ideal level, and gives no indication of how close it is. If the amount by which the amplification level is varied each time is set at a small value, then as shown in  FIG. 9   a  it will take a very long time before the amplification reaches a point near the ideal level and locks in (L 1 ) but after that the overshoot (O 1 ) can be small. If, on the other hand, the amount by which the amplification level is varied each time is set at a large value, then as shown in  FIG. 9   b  the lock in will be quick (L 2 ) but amplification level will repeatedly overshoot the ideal level by a large amount (O 2 ). 
       SUMMARY OF THE INVENTION 
       [0011]    According to the present invention there is provided a multi-rate tracking circuit, comprising: an input for a signal; an output arranged to indicate a current proposed level related to the signal; voting logic connected to the input, arranged to indicate of a direction of change in the current proposed level; a first counter connected to the voting logic, arranged to vary the current proposed level based on the indications received from the voting logic; a second counter, arranged to vary a value based on the indications from the voting logic; wherein the variation of the current proposed level by the first counter is dependent on the value varied by the second counter. 
         [0012]    Advantageously, the first counter is arranged to vary a value based on the indications received from the voting logic, and to vary the current proposed level when the value varied by the first counter crosses a threshold value. 
         [0013]    Preferably, the multi-rate tracking circuit is arranged so that the magnitude of the variation of the value varied by the first counter is dependent upon the value varied by the second counter. Alternatively, the multi-rate tracking circuit is arranged so that the threshold value of the first counter is dependent on the value varied by the second counter. 
         [0014]    Advantageously, the multi-rate tracking circuit is arranged so that the value varied by the second counter is increased in magnitude if the indications from the voting logic cause the proposed level varied by the first counter to be varied consecutively in the same direction. Advantageously, the multi-rate tracking circuit is arranged so that the value varied by the second counter is decreased if the indications from the voting logic cause the proposed level varied by the first counter to be varied consecutively in the different directions. Advantageously, the multi-rate tracking circuit is arranged so that the value varied by the second counter is reset to a default value if the indications from the voting logic cause the proposed level varied by the first counter to be varied consecutively in the different. 
         [0015]    Alternatively, the multi-rate tracking circuit is arranged so that the value varied by the second counter is increased in magnitude if consecutive indications of the same direction are received from the voting logic. Advantageously, the multi-rate tracking circuit is arranged so that the value varied by the second counter is decreased if consecutive indications of different directions are received from the voting logic. Advantageously, the multi-rate tracking circuit is arranged so that the value varied by the second counter is reset to a default value if consecutive indications of different directions are received from the voting logic. 
         [0016]    The second counter may be arranged to vary a second value based on the indications received from the voting logic, and to vary the first value when the second value crosses a threshold value. 
         [0017]    According to the present invention there is further provided a method of tracking a level related to a signal, comprising the steps of: storing a current proposed level, a first value and a second value; determining a desired direction of change in the current proposed level; varying the first value based on the determined direction of change; varying the current proposed level when the first value reaches a threshold value; varying a second value based on the variation of the current proposed level; wherein the magnitude of variation of the first value is dependent upon the second value. 
         [0018]    According to the present invention there is further provided a method of tracking a level related to a signal, comprising the steps of: storing a current proposed level, a first value and a second value; determining a desired direction of change in the current proposed level; varying the first value based on the determined direction of change; varying the current proposed level when the first value reaches a threshold value; varying a second value based on the variation of the current proposed level; wherein the threshold value for the first value is dependent upon the second value. 
         [0019]    Advantageously, the second value is increased when the proposed level is varied consecutively in the same direction. Advantageously, the second value is decreased when the proposed level is varied consecutively in different directions. Advantageously, the second value is reset to a default value when the proposed level is varied consecutively in different directions. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0020]    Examples of the invention will now be described with reference to the accompanying drawings, of which: 
           [0021]      FIG. 1  is a block diagram a receiver circuit, in which the invention may be used; 
           [0022]      FIG. 2  shows the feed forward equaliser and the decision feedback equaliser of the receiver circuit of  FIG. 1 ; 
           [0023]      FIG. 3  is a graph showing the post equalised signal amplitude for exemplary bit patterns; 
           [0024]      FIG. 4  is a diagram of a transmitter; 
           [0025]      FIG. 5   a  shows the response of the receiver to a PRBS transmitted eye-pattern; 
           [0026]      FIG. 5   b  shows the interleaved output of the ADCs of the receiver; 
           [0027]      FIG. 6  is a waveform diagram of a compressed input signal and an amplified input signal; 
           [0028]      FIG. 7  is a graph showing an amplification level approaching an ideal level over time; 
           [0029]      FIG. 8  is a waveform diagram of a digital signal and a corresponding signal showing inter-symbol interference; 
           [0030]      FIG. 9   a  is a graph showing an amplification level approaching an ideal level slowly; 
           [0031]      FIG. 9   b  is a graph showing an amplification level approaching an ideal level rapidly; 
           [0032]      FIG. 9   c  is a graph showing the approach to an ideal level with the invention; 
           [0033]      FIG. 10  is a circuit diagram of a circuit according to the present invention. 
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       [0034]    A key challenge facing designers of high-bandwidth systems such as data-routers and super-computers is the requirement to transfer large amounts of data between ICs—either on the same circuit board or between boards. This data transmission application is called Serialisation-Deserialisation or “SerDes” for short. The present invention is useful in SerDes circuit and indeed was developed for that application. Nonetheless the invention may be used in other applications. 
         [0035]    Analysis of typical backplane channel attenuation (which is around −24 dB) and package losses (−1 to −2 dB) in the presence of crosstalk predict that an un-equalized transceiver provides inadequate performance and that decision feedback equalization (DFE) is needed to achieve error rates of less than 10-17. 
         [0036]    Traditional decision-feedback equalization (DFE) methods for SerDes receivers rely on either modifying, in analogue, the input signal based on the data history [“A 6.25 Gb/s Binary Adaptive DFE with First Post-Cursor tap Cancellation for Serial backplane Communications” R Payne et al ISSCC 2005; “A 6.4 Gb/s CMOS SerDes Core with feed-forward and Decision Feedback Equalization” M. Sorna et al ISSCC 2005; “A 4.8-6.4 Gb/s serial Link for Backplane Applications Using Decision Feedback Equalization” Balan et al IEEE JSSC November 2005.] or on having an adaptive analogue slicing level [“Techniques for High-Speed implementation of Non-linear cancellation” S. Kasturia IEEE Journal on selected areas in Communications. June 1991.] (i.e. the signal level at which the circuit decides whether the signal represents a 1 or a 0). 
         [0037]    A block diagram of a SerDes receiver circuit  1 , which forms part of an integrated circuit, in which the present invention may be used is shown in  FIG. 1 . The invention may nonetheless be used in other applications. 
         [0038]    In the receiver circuit  1  of  FIG. 1  the input data is sampled at the baud-rate, digitized and the equalization and clock &amp; data recovery (CDR) performed using numerical digital processing techniques. This approach results in the superior power/area scaling with process of digital circuitry compared to that of analogue, simplifies production testing, allows straightforward integration of a feed-forward equalizer and provides a flexible design with a configurable number of filter taps in the decision feedback equaliser. The circuit has been implemented in 65 nm CMOS, operating at a rate of 12.5 Gb/s. 
         [0039]    The receiver circuit  1  comprises two baud-rate sampling ADCs (analogue to digital converters)  2  and  3 , a digital 2-tap FFE (feed forward equaliser)  4  and digital 5-tap DFE (decision feedback equaliser)  5  to correct channel impairments. 
         [0040]    The SerDes section of the integrated circuit, which includes the receiver circuit  1  is also provided with a transmitter  40  ( FIG. 4 ), connected to transmit data over a parallel channel to that which the receiver circuit  1  is connected to receive data. The transmitter  40  comprises a 4-tap FIR filter to pre-compensate for channel impairments. In many applications the integrated circuit transmitting data to the receiver circuit  1  uses pre-compensation and in particular a similar transmitter circuit  40 , but in other applications the receiver circuit  1  works without pre-compensation being used at the other end 
         [0041]    The receiver  1  of  FIG. 1  is now described in more detail. The received data is digitized at the baud-rate, typically 1.0 to 12.5 Gb/s, using a pair of interleaved track and hold stages (T/H)  6  and  7  and a respective pair of 23 level (4.5 bit) full-flash ADCs  2  and  3  (i.e. they sample and convert alternate bits of the received analogue data waveform). The two track &amp; hold circuits enable interleaving of the half-rate ADCs and reduce signal related aperture timing errors. The two ADCs, each running at 6.25 Gb/s for 12.5 Gb/s incoming data rate provide baud-rate quantization of the received data. The ADC&#39;s dynamic range is normalized to the full input amplitude using a 7-bit automatic gain control (AGC) circuit  8 . A loss of signal indication is provided by loss of signal unit  9  that detects when the gain control signal provided by the AGC is out-of-range. An optional attenuator is included in the termination block  10 , which receives the signals from the transmission channel, to enable reception of large signals whilst minimizing signal overload. 
         [0042]    The digital samples output from the ADCs  2  and  3  are interleaved and the resulting stream of samples is fed into a custom digital signal processing (DSP) data-path that performs the numerical feed-forward equalization and decision-feedback equalization. This is shown in  FIG. 2 . This comprises a 1 UI delay register  12  connected to receive the stream of samples from the ADCs  2  and  3 . (1 UI is a period of the clock, i.e. the delay between bits.) A tap  13  also feeds the samples from the ADCs to a multiplier  14 , each sample being received by the delay latch  12  and the multiplier  14  at the same time. The multiplier  14  multiplies each sample by a constant weight value (held in a programmable register  15 ), which value is typically 10%. The outputs of the multiplier  14  and the delay register  12  are added together by an adder  16  to provide the output of the FFE  4 . 
         [0043]    The digital FFE/DFE is implemented using standard 65 nm library gates. 
         [0044]    An advantage of applying the equalization digitally is that it is straightforward to include feed-forward equalization as a delay-and-add function without any noise-sensitive analogue delay elements. The FFE tap weight is selected before use to compensate for pre-cursor ISI and can be bypassed to reduce latency. Whilst many standards require pre-cursor de-emphasis at the transmitter, inclusion at the receiver allows improved bit error rate (BER) performance with existing legacy transmitters. 
         [0045]    The DFE  5  uses an unrolled non-linear cancellation method [“Techniques for High-Speed implementation of Non-linear cancellation” S. Kasturia IEEE Journal on selected areas in Communications. June 1991]. The data output (i.e. the 1s and 0s originally transmitted) is the result of a magnitude comparison between the output of the FFE  4  and a slicer-level dynamically selected from a set stored in a set  17  of pre-programmed registers. The values are determined by a control circuit (not shown in  FIG. 1 ) from the waveforms of test patterns sent during a setup phase of operation. The magnitude comparison is performed by a magnitude comparator  18  connected to receive the output of the FFE  4  and the selected slicer-level; it outputs a 1 if the former is higher than the latter and a 0 if it is lower or equal, thereby forming the output of the DFE  5 . 
         [0046]    The slicer-level is selected from one of 2n possible options depending on the previous n bits of data history. The history of the bits produced by the magnitude comparator  18  is recorded by a shift register  19  which is connected to shift them in. The parallel output of the shift register is connected to the select input of a multiplexer  20  whose data inputs are connected to the outputs of respective ones of the set  17  of registers holding the possible slicer-levels. 
         [0047]    Unrolled tap adaption is performed using a least mean square (LMS) method where the optimum slicing level is defined to be the average of the two possible symbol amplitudes (+/−1) when proceeded by identical history bits. (For symmetry the symbols on the channel for the bit values 1 and 0 are given the values +1 and −1). 
         [0048]    Although 5-taps of DFE were chosen for this implementation, this parameter is easily scaleable and performance can be traded-off against power consumption and die area. In addition, the digital equalizer is testable using standard ATPG (automatic test pattern generation) and circular built-in-self-test approaches. 
         [0049]    The chosen clock recovery approach uses a Muller-Mueller approach [“Timing recovery in Digital Synchronous Data Receivers” Mueller and Muller IEEE Transactions on Communications May 1976.] where the timing function adapts the T/H sample position to the point where the calculated pre-cursor inter-symbol interference (ISI) or h(−1) is zero, an example being given in  FIG. 3 . The two curves show the post-equalized response for 010 and 011 data sequences respectively. The intersection  30  at 3440 ps occurs when the sample of the second bit is independent of the third bit—that is, h(−1)=0. This position can be detected by comparing the post-equalized symbol amplitude with the theoretical amplitude h(0) and using the difference to update the CDR&#39;s phase-interpolator. 
         [0050]    A block diagram of the transmitter is shown in  FIG. 4 , which is implemented using CML techniques. The data to be transmitted (received at terminal  41 ) is sequentially delayed by three 1 UI delay registers  42 ,  43  the  44  connected in series. They produce, via the four taps before and after each delay, a nibble-wide word containing the pre-cursor, cursor and two post-cursor components. In fact to ease timing closure the data is sent to the transmitter from the digital part of the circuit that supplies the data in blocks of 4 nibbles (16 bits in parallel), the blocks being sent at a rate of 3.125/s. Each nibble is a frame of four bits of the bitstream offset by one bit from the next so the nibbles overlap and represent the data redundantly. A multiplexer then selects one of the nibbles, switching between them at a rate of 12.5×109/s, and presents that in parallel to the four taps, thereby making the bitstream appear to advance along the taps. 
         [0051]    A 4-tap FIR output waveform is obtained from simple current summing of the time-delayed contributions. This is done with differential amplifiers  45  to  48 , each having its inputs connected to a respective one of the taps and having its differential output connected to a common differential output  49 . Although shown as four differential amplifiers the circuit is implemented as one differential amplifier with four inputs, which minimizes return-loss. The relative amplitude of each contribution is weighted to allow the FIR coefficients to be optimized for a given circuit (e.g. a backplane) and minimize the overall residual ISI. The weights are determined empirically either for a typical example of a particular backplane or once a backplane is populated and are stored in registers  50  to  53 . The weights respectively control the controllable driving current sources  54  to  57  of the differential amplifiers  45  to  48  to scale their output current accordingly. Respective pull-up resistors  58  and  59  are connected to the two terminals of the differential output  49 . 
         [0052]    A PLL is used to generate low-jitter reference clocks for the transmitter and receiver to meet standards [“OIF-CEI-02.0—Common Electrical I/O (CEI)—Electrical and Jitter Interoperability agreements for 6 G+ bps and 11 G+ bps I/O”. Optical Internetworking Forum, February 2005; “IEEE Draft 802.3ap/Draft 3.0—Amendment: Electrical Ethernet Operation over Electrical Backplanes” IEEE July 2006.]. Most integrated circuits will have more than one receiver  1  and the PLL is shared between them with each receiver having a phase interpolator to set the phase to that of incoming data. 
         [0053]    The PLL uses a ring oscillator to produce four clock-phases at a quarter of the line data-rate. The lower speed clocks allow power efficient clock distribution using CMOS logic levels, but need duty-cycle and quadrature correction at the point of use. The 3.125 GHz clocks are frequency doubled (XOR function) to provide the 6.25 GHz clock for the T/H &amp; ADC. The transmitter uses the four separate 3.125 GHz phases, but they require accurate alignment to meet jitter specifications of 0.15UI p-p R.J. and 0.15UI p-p D.J. 
         [0054]    The system described has been fabricated using a 65 nm CMOS process and has been shown to provide error-free operation at 12.5 Gb/s over short channels (two 11 mm package traces, 30 cm low-loss PCB and two connectors). A legacy channel with −24 dB of attenuation at 3.75 GHz supports error free operation at 7.5 Gb/s. 
         [0055]      FIG. 5   a  shows a 12.5 Gb/s 27-1 pseudo random bit stream (PRBS) transmitted eye-pattern with 20% de-emphasis on the first post-cursor. The receiver includes, for test purposes, a PRBS data verifier  66 , which confirms that the test pattern has been received. The differential peak-to-peak (pp) amplitude is 700 mV (200 mV/div).  FIG. 5   b  shows the ADC output when a 6.25 GHz sine-wave is sampled and the phase between the sine-wave and receiver is incremented using a programmable delay-line. The measured codes are within +/−1 lsb (least significant bit) of the expected values. This level of performance ensures robust operation over a wide range of cables, green-field and legacy channels. The worst-case power of a single TX/RX pair, or “lane” is 330 mW and the total exemplary macro area is 0.45 mm2 per lane (allowing for the PLL being shared by four TX/RX lanes. 
         [0056]    An example of a circuit according to the present invention is shown in  FIG. 10 . A voting logic circuit  1000  takes as input a signal I, the compressed signal which is being compensated, and also the current level of amplification L. The voting logic  1000  has outputs INC and DEC, which are used as input for an ‘N’ counter (or accumulator)  1001 . The ‘N’ counter  1001  has outputs INC and DEC, which are used as input for a ‘step’ counter  1003  and a register  1002 . The output of the register  1002  is L, which is used as an input of the voting logic  1000  as mentioned above. The ‘step’ counter  1003  has outputs INC and RESET, which are used as inputs for a register  1004 . The output of the register  1004  is used as an input for the ‘N’ counter  1001 . 
         [0057]    In operation, the voting logic  1000  assesses the input signal I and current level of amplification L, and indicates via its outputs INC and DEC whether the amplification level needs to be incremented (because the current amplification is too low) or decremented (because the current amplification is too high) respectively. 
         [0058]    The ‘N’ counter  1001  operates using a preset threshold value N, which does not change during operation of the circuit. The ‘N’ counter  1001  stores a value M, which is incremented or decremented according to the inputs from the voting logic  1000 . M is incremented or decremented by S, the value of which is set by the register  1004 . If M is incremented to above the threshold value N then the ‘N’ counter  1001  sends an increment signal down its INC output to the register  1002 , and M is reset to N/2 (half of N). If M is decremented to below zero then the ‘N’ counter  1001  sends a decrement signal down its DEC output to the register  1002 , and M is reset to N/2 (half of N). 
         [0059]    The register  1002  stores the current level L of amplification, which it increments or decrements when it receives a corresponding signal from the ‘N’ counter  1001 . 
         [0060]    The ‘step’ counter  1003  also operates using a preset threshold value P, which again does not change during operation of the circuit. The ‘step’ counter stores a value C. Suppose the previous signal received from the ‘N’ counter  1001  was an increment signal. If the ‘step’ counter  1003  receives another increment signal, then C is incremented by 1. Similarly, if the previous signal received from the ‘N’ counter  1001  was a decrement signal, and the ‘step’ counter  1003  receives another decrement signal, then C is again incremented by one. If C reaches the threshold value P then the ‘step’ counter sends an increment signal down its INC output to the register  1004 , and C is reset to zero. 
         [0061]    However, if the ‘step’ counter  1003  receives an increment signal when it had previously received a decrement signal, or a decrement signal when it had previously received an increment signal, then it sends a reset signal down its RESET output to the register  1004 , and C is again reset to zero. 
         [0062]    The register  1004  stores a value S. If the register  1004  receives an increment signal from the ‘step’ counter  1003  then it increments S. If, on the other hand, it receives a reset signal from the ‘step’ counter  1003  then it resets S to one. 
         [0063]    To see how the circuit works, suppose that initially the value stored in the register  1002  (that is the amplification level L) is L 0 , and the ideal amplification level is L*. This is the situation shown in  FIG. 7 . Initially the value M stored in the ‘N’ counter  1001  is set at N/2, the value C stored in the ‘step’ counter  1003  is zero, and the value S stored in the register  1004  is one. 
         [0064]    As the current amplification L is below the ideal amplification L*, the voting logic  1000  will output repeated increment signals. This will repeatedly increment the value M stored in the ‘N’ counter  1001  by one (as S is one). When M reaches the threshold value N the value L in the register  1002  is incremented, M is reset to N/2, and the cycle repeats. (The voting logic  1000  may due for example to inter-symbol interference output a decrement signal, but these will occur rarely enough that the ‘N’ counter  1001  will always eventually reach the threshold value N.) 
         [0065]    At the same time as the above is occurring, each time the ‘N’ counter  1001  reaches the threshold value N the value C stored in the ‘step’ counter  1003  is incremented. When C reaches the threshold value P, the value S is incremented. This causes the value M in the ‘N’ counter  1001  to be incremented more quickly, which in turn causes the amplification level L to be incremented more quickly (as it takes fewer increments from the voting logic  1000  before the value M reaches the threshold value N). Thus it can be seen that the time frame by which the compression compensation circuit operates is reduced as the value S stored in the register  1004  increases. As more consecutive increment signals are output by the ‘N’ counter  1001  the value of S will continue increasing, causing even greater reduction in the time frame by which the compression compensation circuit operates. 
         [0066]    However, once the amplification level L exceeds the ideal amplification level L*, the voting logic  1000  will being to output decrement signals. This will decrement the value M stored in the ‘N’ counter  1001 . Eventually the value M will reach the threshold value of 0, causing the value L in the register  1002  to be decremented. However, as the decrement signal follows an increment signal, it will also cause the ‘step’ counter  1003  to send a reset signal to the register  1004 , which resets S to one. This causes the value M stored in the ‘N’ counter  1001  to be incremented or decremented by only one, thus returning the operation of the compression compensation circuit to its longest time frame of operation. 
         [0067]    It can thus be seen that the circuit of  FIG. 10  is able to approach an ideal amplification level rapidly when it is far away, but once the ideal level has been reached the rate of change of the amplification level is reduced, preventing the circuit from continually overshooting the ideal level. This is illustrated in  FIG. 9(   c ) where a quick lock (L 2 ) is provided along side a low overshoot (O 1 ) 
         [0068]    It can be seen that the circuit will operate in a similar way when the ideal amplification level L* is lower than the initial amplification level L 0 . 
         [0069]    Another example of a circuit according to the present invention is shown in  FIG. 11 . The circuit is the same as that of the previous example, except that the ‘N’ counter  1001  has been replaced by and alternate ‘N’ counter  1010 . In this case, the alternate ‘N’ counter  1010  takes as threshold values zero and N−S, outputting a decrement or increment signal when its internal value M crosses those respective thresholds. When a threshold is crossed, M is reset to (N−S)\2. It can be seen that the circuit will operate in much the same way as the circuit of the previous example, except that the time frame of operation of the circuit will be reduced at a slower rate. 
         [0070]    It will be appreciated that various other changes to the circuits described could be made which would still be covered by the present invention. For example, the value S stored by the register  1004  could be incremented exponentially (taking the values 1, 2, 4, 8 and so on) rather than by one each time. In a further example, the threshold value P of the ‘step’ counter  1003  could be dependent on the value S stored by the register  1004 , for example being reduced as S increases. In a further example, there could be no register  1004 , and the increment size S used by the ‘N’ counter  1001  (or alternative ‘N’ counter  1010 ) could be the number of consecutive increment or decrement signals provided by the voting logic  1000 . In a further example, the value S stored by the register  1004  could be incremented when consecutive increment or decrement signals are received from the voting logic  1000  rather than the ‘N’ counter  1001 . In a further example, the value S stored by the register  1004  could be incremented when consecutive increment or decrement signals are received from the ‘N’ counter  1001  as described above; however, instead of being reset when there is a change in signal from the ‘N’ counter  1001 , such a change could have no effect, and the value S could be decremented at regular periods instead. Such a circuit would thus gradually increase its time frame of operation once the ideal level had been reached. 
         [0071]    Although this circuit has been described in the context of a compression compensation circuit, it will be appreciated that it is relevant to any situation in which the finding and maintaining of a particular level is required. The level need not be used to vary the signal itself, but could relate to finding a characteristic of the signal, such as its frequency. For example, if the circuit was a phase-locked loop being used to track the frequency of a signal, the voting logic would indicate whether the phase of the circuit was too early or too late.