Abstract:
An apparatus and method make use of a single shunt and two or more instrumentation amplifiers, switchably measuring voltages at the shunt. This permits current measurement. At times each instrumentation amplifier has its input shorted, which permits zeroing out many sources of offset in the signal path of that amplifier. Dynamic range is several orders of magnitude better than known current measurement approaches, permitting coulometry.

Description:
Application Ser. No. 13/580,377 filed Dec. 7, 2011 is incorporated herein by reference for all purposes. 
     BACKGROUND 
     It is not easy to measure the flow of electricity accurately over a wide dynamic range while dissipating very little waste heat. 
     Whenever electrical energy is used, it is desirable to measure the quantity of energy used, both per unit of time (e.g. power), and over a specific amount of time (for example, energy used per month). In a residential environment, an electric power meter is a familiar fixture; it allows the power-providing companies to charge their customers for the energy used. 
     In the most basic terms, the instantaneous power is a product of the voltage applied to, and the current flowing through, the load. An integration of this product (power) over a specific time interval yields the total consumption of the electrical energy within that time interval. 
     Accurate measurement of the load current is thus an important part of the apparatus that measure the consumption of electrical energy. 
     It will be helpful to review the present state of the art in current measurements, considering numerous examples of the devices capable of current measurement. In broad terms, they all fall into two categories, namely:
         direct measurement of the current as manifested by a voltage drop across a sense resistor (oftentimes called a resistive shunt), and   indirect measurement of the flowing current by assessment of the magnetic field that is created around the conducting wire that is carrying the current.       

     Current-measuring apparatus further differ in their ability to measure DC (direct current) or AC (alternating current), and also differ in the total range of frequencies that can be processed with a high degree of accuracy. The apparatus that can accommodate large currents also differ in their ability to measure small currents accurately. 
     Typical current sensors in industrial, commercial, and household environments furnish the AC measurements; the parameter of interest is usually the peak-to-peak or Root-Mean-Square (RMS) value of the current, averaged across at least a single cycle of the AC frequency; in such measurements the DC component of the current is assumed to be exactly zero. The ultimate use of these measurements is for an overall consumption of energy, and the sensor is made to be quite accurate when a large level of current is flowing. Errors resulting from inadequate precision of measurements for small levels of current are simply ignored, as they contribute only minimally to the total energy assessment. Stated differently, when one is measuring large currents to assess overall energy consumptions, one does not mind that small currents are not measured accurately. 
     A moment&#39;s reflection, however, will reveal that for operations with DC power, and specifically for DC-power systems that use a battery, the current measurement needs are quite different. It is not uncommon for a DC-supplied system to spend most of its time operating at a very low power level, and to consume the full-rated energy only for relatively brief intervals. In such a system, the current-sensing apparatus must be able to faithfully measure the current across widely different levels, while at the same time maintaining at most a very small DC offset error. In many DC power systems that use a battery, it is desired to monitor the SOC (state of charge) of the battery which requires continuous and uninterrupted current measurements so that integrations may be carried out, making coulometry possible. 
     Several contradictory constraints apply to such a current sensor, for example:
         The output signal should accurately represent the measured current. In technical terms, the sensitivity or conversion gain between the measured current and output signal should be accurate and stable.   The readings for times when no current is flowing should be as close as possible to zero. In technical terms, the systematic offset of the measurement should be zero.   The output signal should be available continuously, with quick response time to the changes in the measured current.   The current sensor should be insensitive to interfering RF (radio frequency) and EM (electromagnetic) fields, because such fields are naturally present in any environment where electrical energy is used, and are typically generated by the very same apparatus that use the electric energy. This requirement usually dictates the use of so-called RFI (RF interference) filtering, in order to prevent the interfering signals from affecting the measurements.   The power dissipation in the sensing element should be minimal.   The energy consumed by the current measurement circuit should be small; this has an increased importance for a system that a operates from a battery, and the actual goal of making the current measurements is for reduction of the energy consumption and/or estimation of the remaining energy in the battery. Having said this, in many applications it is acceptable to permit the energy used in the current measurement circuit to increase somewhat when large currents are flowing, that is, when large amounts of energy are being consumed. In other words, it is desirable to have low energy consumption in relative terms, where the power consumption of the measurement circuit will not appreciably affect the total energy value.       

     The most common method of current sensing is to pass the current through a resistor (a current shunt) and to measure the resulting voltage drop, which develops according to Ohm&#39;s law. A current sensor circuit based on this principle is illustrated in  FIG. 1 . input terminals  1  allow connection of the current shunt  2  into the circuit where current is to be measured. Instrumentation amplifier (IA)  6  senses voltage across the current shunt  2 , and provides (voltage) output on terminals  10 . Pick-up points  3  on the current shunt  2  are located following the principle of “Kelvin sensing” that reduces errors associated with resistance of the sense connections and wires. With such an arrangement, the conductors that do the sensing have almost no current, and so there is very little IR (current times resistance) drop in those conductors. RFI filter  4  is utilized for reduction of effects due to interfering RF signals from surrounding machinery and environment. This part of the circuit may have various configurations, and may consist of resistive, capacitive, and inductive components, as well as semiconductor transient protectors. The particular configuration  4  shown in  FIG. 1  is exemplary but there are many possible arrangements which might be employed. 
       FIG. 1  shows an instrumentation amplifier  6  modeled as containing an ideal amplifier  7 . The ideal amplifier  7  is imagined to have infinite input impedance, zero output impedance, linear gain, and zero offsets or drift, all across some defined dynamic range of inputs and from DC to at least some frequency defining some frequency response. Also shown in  FIG. 1  are modeled sources of error  5 ,  8 , and  9  that make the performance less than ideal when compared to the requirements above. The modeled voltage sensing error sources  5  appear in series with the proper signal, and introduce unpredictable offset errors at the inputs of the IA  6 . These errors can be further modeled as illustrated in  FIG. 2 . 
     Turning to  FIG. 2 , items comprising a string of dissimilar materials  13 ,  14 ,  15 ,  17 ,  18 , and  20  are joined by solder  16 . Exemplary items may be a Kelvin connection lead  13  from the current shunt  2 , a surface-mount component  15  such as a resistor, an IC (integrated circuit) lead  17 , a lead-to-die bonding wire  18 , and IC die metallizations  20  at the IC die  19 . A printed circuit (PC) board  12  is composed in part of a substrate material such as FR-4. A modeled temperature gradient  21  is shown, with a higher temperature at one end of this string and a lower temperature at the other end of this string. As mentioned below, anisotropies and inhomogeneities of temperature may easily develop to be more complicated than a mere gradient as shown in  FIG. 2 . As will be discussed below in connection with the invention, approaches discussed below have some prospect of measuring and correcting errors due to temperature variations that are more complicated than mere gradients. 
     The materials depicted in  FIG. 2  make up the connections between the current shunt  2  and the actual input terminals of the IA  6  (which are part of the die metallization  20 ). Redrawn in a more recognizable form at the lower part of  FIG. 2 , it will be appreciated that these connections are in fact thermocouples, each capable of producing a voltage when a temperature difference (temperature gradient  21 ) exists between the extremes of the circuit at points  13  and  20 , or inside of the string of the serially connected elements. This error voltage could potentially be on the order of several millivolts (mV). The most common temperature-induced errors may be because of gradients between 13 and 20, but as will be appreciated, any anisotropies or inhomogeneities in temperature can likewise give rise to error voltages. 
     Returning to  FIG. 1 , the error source  8  is a modeled offset error in the IA  6 , and the error source  9  is a modeled noise error in the IA  6 . 
     As will be appreciated, the circuit is designed utilizing differential signals, and the designer will hope that this will compensate most of the thermocouple-induced errors, because such errors are to some extent common-mode in nature (affecting each line of the two differential inputs in somewhat the same way). But any non-identical temperature distributions as between the two paths can give rise to errors which are not automatically compensated by the use of differential sensing. 
     Because there are error sources  5  in the differential sensing lines, the designer of a system such as is depicted in  FIG. 1  will often not go to the trouble to select a module or integrated circuit for IA  6  that has a particularly low offset voltage  8 . 
     The circuits similar to  FIG. 1 , or derivatives thereof, are generally used for measuring currents up to perhaps 10 A (Amperes). The designer of the current sensing system will typically select a shunt value such that the voltage drop across the sensing resistor  2  is 100 mV or more, so that this voltage will swamp or overwhelm the error sources just described. For any given maximum rated current (that is, for some particular real-life application) the resistance of the shunt  2  will be chosen so that the voltage developed across it is within some range of voltages. This leads to a situation where the heat dissipated in the sense resistor  2  grows at least linearly with the rated maximum current for which the particular circuit is being designed. 
     The physical size of the sense resistor  2  can thus get to be a problem, as can be the need to providing adequate cooling of the sense resistor. For this reason, for higher currents a different class of current sensing devices is used, typically utilizing an indirect method of current sensing by assessment of the magnetic field that is created around the conducting wire which is carrying the current. 
     The so-called Hall-effect apparatus is common for current measurements in excess of several Amperes. A Hall-effect device is able to produce a (differential) voltage signal when a continuous (supply) current is sent through the device, and a magnetic field is present that is perpendicular to the flow of that supply current. The voltage signal in the Hall-effect device is linearly proportional to both the supply current and the magnetic field, within limitations of power dissipation resulting from the supply current, and some additional anomalous effects. The current that needs to be measured generates the magnetic field acting on the device. 
     The sensitivity of the Hall-effect device to the magnetic field depends, among many things, for example upon mechanical dimensions of the sensing element, material composition and uniformity, attachment of the electrodes, stability and accuracy of the supply current, and construction of the magnetic core that concentrates the desired magnetic field and rejects interfering magnetic fields (of which the Earth&#39;s magnetic field is just a single example). 
     Typically every single manufactured Hall-effect device would need an individual calibration in order to ascertain the actual device sensitivity (which further depends on the particular circuit arrangement and on the design of the magnetic path elements). 
     The native offset of the unaided Hall-effect device is nonzero. For this reason various auto-zeroing schemes have been utilized in the prior art. One approach is to arrange for the supply current to the sensing element to be AC rather than DC. Within such an approach, differing schemes make use of various shapes or waveforms of the AC current, for example sine waves or square waves or even more complex shapes. The AC signal that is the output of the Hall-effect sensing element is further processed by a synchronous detector. 
     In a system where the Hall-effect sensor is employed and where the current sensed is AC, it usually turns out to be necessary to average or filter the sensed signal over several cycles of the AC excitation. This means that there is always some latency between a current event of interest and the detection of such an event after the averaging or filtering has taken place. The latency cannot be reduced to zero. 
     A further potential difficulty with such magnetically coupled measurements (particularly where a DC current is converted to AC for purposes of Hall-effect sensing) is that during the zero crossings of the excitation voltage (that is, near the zero values for the sine-wave excitation), the Hall-effect device is altogether insensitive to the magnetic field, and simply discards any information for the duration of the zero-crossing transitions. 
     Yet another variation of the Hall-effect devices makes use of an active feedback loop that tries to zero-out the total magnetic field acting on the sensing element, thus reducing possible non-linearities in the sensing element (since in this case, the Hall-effect sensing element only needs to indicate if the magnetic field is smaller or higher than zero, and does not need to supply the actual value). A winding on the magnetic path elements creates a magnetic field that is opposite to the field from the measured current. The opposing winding is normally constructed with many turns (typically, with several thousands of turns), so as to minimize the current that needs to be injected into the winding in order to zero out the total magnetic field. The servo loop that drives the zeroing-out winding will of course take some nonzero time to settle and to respond to perturbations. 
     It will be appreciated that in addition to slow time response, the Hall-effect devices utilizing the active feedback will require additional operating energy in order to supply the feedback circuitry and winding. These factors make the Hall effect sensing less than ideal, particularly for a battery-powered system. 
     Other current measurement approaches have been devised that indirectly sense current by assessment of the magnetic field that is created around the conducting wire that is carrying the current. These are based on a so-called magnetoresistance (MR) effect, including variations called Giant Magnetoresistance (GMR) and Colossal Magnetoresistance (CMR). These devices rely on resistance changes within the sensing material due to the magnetic field to which the sensing material is subjected. 
     The error sources mentioned above with respect to shunt-type current measurements present themselves, mutatis mutandis, with the Hall-effect sensing approaches just discussed, and also present themselves with the apparatus based on the MR-effect. For the MR-effect approaches, the notably worst performance is in respect to the zero-current offset for the MR-effect based measurements. 
     It would be very desirable if approaches could be found for current measurement, particularly DC measurement, which would be accurate at low currents as well as high currents, and which would be continuously available. It would be desirable if most offset error sources could be zeroed out, leading to measurements accurate enough for coulometry. It would be desirable if the approaches could dissipate very little waste heat in the sensing apparatus, and if the measurements could simultaneously:
         be low enough in latency to permit a quick-response electronic “fuse” capability, and yet   be sufficiently filtered against RFI so as to permit filtering out such interference.       

     It would further be desirable if some or all of these results could be achieved economically, without the need for unnecessarily expensive parts such as unnecessarily expensive semiconductor switches and the like. 
     SUMMARY OF THE INVENTION 
     An apparatus and method make use of a single shunt and two or more instrumentation amplifiers, switchably measuring voltages at the shunt. Dynamic range is several orders of magnitude better than known current measurement approaches, permitting coulometry. 
     The current invention teaches a circuit and an algorithm (for the control part of the circuit, and for combined operations of the analog and digital parts of the circuit). 
     This circuit and the algorithm empower the creation of a current sensor that is accurate at much larger range of measured currents than in the prior art; it provides several-orders-of-magnitude dynamic range improvement. Compared with the prior-art implementations, the current invention reduces the waste of energy in the sensing element to near zero. The energy consumption of the circuit itself can be reduced to near zero levels under conditions of low reporting rates for the measured current and/or accumulated charge. Due to very low measurement offset for the DC current, an accumulated charge value is quite accurate, even under conditions of a load current that is many times smaller than the maximum rated load current. 
     The high accuracy of the current sensor is ensured due to continuous assessment of most of the interfering voltage errors, and subtracting these errors from the actual measured value. 
     The output of the circuit according to the current invention is continuous, if such an operating mode is desired; this is in spite of the fact that all circuits in the sensor alternate between the actual measurement and the calibration functionality (in order to measure and zero-out the errors). 
     The ability to provide a continuous output is important if fast-acting electronic fuse functionality is desired. The ability to provide a continuous output is important as well if it is desired to integrate the measured current so as to achieve coulometry. 
     The actual frequency, with which the calibration cycles are performed, can be fixed or it can depend on the changes in the temperature or temperature gradients that are the dominant causes of the errors. In other words, the calibration cycles can be performed more frequently if the temperatures and/or temperature gradients change rapidly. 
    
    
     
       DESCRIPTION OF THE DRAWING 
       The invention will be described with respect to a drawing in several figures, of which: 
         FIG. 1  depicts the prior art circuit, 
         FIG. 2  illustrates generation of voltage errors, 
         FIG. 3  describes one current measurement approach according to the invention, 
         FIG. 4  shows the algorithm for activations of FETs  73 / 74  and Mux  48  signal in  FIG. 3 , 
         FIG. 5  presents a physical implementation that reduces uncompensated errors, 
         FIG. 6  demonstrates an alternative embodiment with two RFI filters in each channel, 
         FIG. 7  shows the apparatus in a second embodiment according to the invention, with a single shunt and a two-FET switching circuit, and 
         FIG. 8  shows an alternative embodiment according to the invention, with a two-FET switching circuit and two RFI filters in each channel. 
     
    
    
     DETAILED DESCRIPTION 
     One current sensor according to the invention is shown in  FIG. 3 . There is a current shunt  72  providing a voltage signal proportional to the current flowing in the shunt  72  to resistors  75   a - 75   d , which are in turn connected to field effect transistor (FET) switches  73 / 74 . Further, current sense signals pass via RFI filters  37  and  38 , and then get amplified by instrumentation amplifiers (IAs)  41  and  42 . An analog selector switch  43  delivers the signal to output terminals  44 ; this signal can be either the output from IA  41 , or IA  42 . 
     Under control of the circuit  51 , and via FET Drivers  45 , the FETs  73 / 74  are turned either fully on or off, as required for the execution of the Algorithm detailed in  FIG. 4 . 
     The action of FETs  73 / 74  can short-circuit the signal voltage input to RFI filters  37 / 38 ; offset voltage errors associated with the RFI filters and the whole amplification chain  37 / 39 / 41  or  38 / 40 / 42  can then be calibrated out. 
     At any point in time, there will be at least a single FET turned off from the pair  73 / 74 . 
     Also, at any moment in time, there is at least a single RFI filter and an amplifier that provides output to the Output Terminals, via IA  41  or  42 . 
     When it is desired to measure the voltage sensing error, the current sense signal is removed from one of the measurement channels by activation of the shunt FET (e.g.  73  or  74 ); the current sense signal is reduced in proportion to the ratio of resistors  75   a - 75   d  and on-resistance (so-called Rds-on) of the switches  73 / 74 . 
     When current sense signal is removed from one of the measurement channels, the un-energized channel (via IA  41  or IA  42 ) measures the voltage errors, and Control Circuit  51  adjusts the output of Digital-to-Analog (D/A) converter  46 , until the output of the channel being calibrated is equal to zero. 
     It will be appreciated that if continuous analog output signal is not required, and the only output from the current sensor circuit will be via the digital communication interface  50 , then analog switch  43  as well as D/A  46  are not required, and removal of the voltage error can be done entirely in digital circuitry inside of the control circuit  51 , by simply subtracting the measured error from the total signal measured by the analog-to-digital (A/D) converters  47 . 
     The operations described above with respect to shunting the current sense (voltage) signal almost at the source differ greatly from the typical prior-art approach of reducing the voltage errors by auto zeroing the voltage offsets inside of the amplification chain; the prior-art approach cannot remove the errors associated with components located between the sense points and the inputs of the amplifier. Also, the prior-art approach is not able to compensate the errors resulting from lead-to-die bonding inside of the solid-state integrated circuits. 
     However, due to a much-simplified switch matrix (as compared to one previous circuit) the errors linked with Kelvin-sensing connections of the leads of the current shunt  72 , as well as error voltages associated with resistors  75   a - 75   d  and switches  73 / 74  are not automatically compensated for. 
     On the other hand, there are only two MOSFET switches (as opposed to four in one previous circuit), and these switches  73  and  74  can be much smaller, and are only required to be able to carry a very small current, and thus are much less expensive than the units described in one previous circuit. The on resistance (Rds-on) parameter of switches  73  and  74  can be tailored to the values of resistors  75   a - 75   d , and indeed the ratio between the resistances of resistors  75   a - 75   d  and Rds-on for the switches  73 / 74  define the ultimate accuracy and dynamic range of the circuit. 
     In order to reduce the uncompensated errors associated with resistors  75   a - 75   d  and switches  73 / 74 , a specific physical construction is utilized, as presented in  FIG. 5 . 
     Current shunt  72  ( FIG. 5 ) is constructed from copper utilizing common Printed Circuit Board (PCB) techniques. This current shunt does not have explicit sensing leads; sense lines  78  going to resistors  75   a - 75   d  are continuous in respect to the body of the current shunt  72 . 
     In other words, there is no junction of dissimilar materials between the sensing leads on the current shunt and copper traces connecting to the rest of the circuit, thus no thermoelectric voltages can be generated. 
     The substrate of this PCB  70  may be a standard material called FR-4, or any other suitable for the required operating temperature range. 
     Components  75   a - 75   d ,  73 , and  74  are closely grouped together in a small area of the PCB adjacent to the current shunt  72 . 
     Metallization on the leads of resistors  75   a - 75   d  and switches  73 / 74  in combination with copper traces on the PCB (and with interloping solder) indeed create pairs of dissimilar materials that can lead to generation of thermoelectric error voltages; however, if the temperature of all such pairs of dissimilar materials is the same, then the resulting error voltages are very small or zero. 
     Furthermore, even if there is some temperature misbalance within the grouped components  75   a - 75   d ,  73 , and  74 , the thermoelectric voltage across each of two resistors  75   a - 75   d  that are connected to the same MOSFET switch  73  or  74  will have the alike value, and will cancel each other due to the differential-sensing circuit that follows it. 
     Moreover, a slot (air gap)  77  is cut into the PCB; this prevents the heat flux from escaping the area of the grouped components through heat conduction via the PCB substrate, effectively enforcing the same temperature across the whole area of the grouped components  75   a - 75   d ,  73 , and  74 . 
     A careful observer will recognize that all interconnections between grouped components  75   a - 75   d ,  73 , and  74  are made with traces that are located on the same side of the PCB, without intervening vias (copperplated holes that connect traces on different layer of PCB, e.g. top and bottom). This prevents errors (thermoelectric and, possibly, electro-mechanical) that can result from particular methods of construction of the vias, which vary widely between PCB manufacturers. 
     On the other hand, the connections to the rest of the circuit are made through vias  76 . While these connections to the rest of the circuit will pass through the areas with uneven temperatures and will indeed produce thermoelectric error voltages, their effects will be totally negated by the calibration action of the circuit. 
     Going back to  FIG. 3 , the two amplification channels are present in order to allow a single channel to provide the continuous readout of the current, while the other channel is calibrated. Calibration is performed when the current sense signal has been short-circuited by one of the MOSFET switches  73  or  74 , and RFI filter  37  or  38  with corresponding IA  41  or  42  have settled to a final value that represents zero value of measured current; then auto zeroing action is executed by Control Circuit  51  (e.g. adjustment of the D/A to null-out the output of the appropriate IA  41  or  42 ). Once calibration is finished, the inactive channel is switched back into service, by opening one of the MOSFET switches  73  or  74 . After RFI filter  37  or  38  have settled to a new value that is representative of the current being measured, the Analog Switch  43  is used to switch the Output terminals  44  to this newly calibrated channel. Thus, there are times in this circuit when both channels have output representative of the current being measured. During such times, the Control Circuit  51  can cross correlate the two signals; total accuracy of the Current Sensor can be improved by averaging the two values, and problems with the circuit can be discovered and reported if the readings from two channels start to diverge. 
     The user may elect to connect the local circuit&#39;s common potential (e.g. ground) to the middle of the current shunt  72  through the connection  79 . This will give an improved performance for the RFI filters due to symmetrical differential nature of the RFI filters and the following circuit. Furthermore, the ground connection  79  to the shunt provides a reference voltage defining a fixed level at the sources of the MOSFETs  73 ,  74 . This permits the system designer to select the output voltage at drivers  45  to fully turn the MOSFETs  73 ,  74  on or off. 
     It may be instructive to say a little more about the resistors  75   a - 75   d . The value of these resistors cannot be permitted to be too large because mismatch in the bias current of the AI will create systematic offset error due to I*R losses. By way of perspective, a 1 nA bias mismatch (which is a possible value for an IA) across a 100 Ohm resistor will create an offset of 100 n V. 
     On the other hand, the same sorts of errors from resistors in block  37  do not affect the output of an IA, since those errors are “after” the MOSFETs  73 ,  74  in the signal path, and these errors are zeroed by the calibration/autozero action. This permits the resistors in block  37  to have relatively high values if desired. This may be helpful for example if it is desired to use reasonably small capacitors in the block  37  while nonetheless achieving meaningful time constants in the filters. 
     Returning to resistors  75   a - 75   d , while it would be a problem if the resistors were too large in value, it is also the case that resistors  75   a - 75   d  cannot be permitted to be too small, since we want to attenuate the signal across shunt  72  as much as possible when autozero is being performed. In other words—the ratio between the value of resistors  75   a - 75   d  and the “on” resistance of MOSFETs  73 ,  74  should be as high as possible. So consider for example the use of 100-Ohm resistors  75   a - 75   d , with MOSFETs  73 ,  74  that have 10 milliOhms on resistance (which is quite low). In this example, the largest error coming from this noninfinite attenuation when a MOSFET is on is on the order of a full-scale signal divided by 10 k, or only 80 dB of signal-to-noise. 
     The circuit&#39;s operating algorithm will now be described referring to  FIG. 4 . 
     An arbitrarily selected point in time  81  is chosen as the beginning of the algorithmic cycle; at this time both MOSFET switches  73  and  74  are off, and both channels are providing the current signal; however Mux  48  signal has just selected the output of IA  41  to be passed to output terminals  44 . 
     At point  82  the switch  74  is turned on, and the current sense signal (voltage) at RFI filter  38  is short-circuited by MOSFET switch  74 . The voltage at the input terminals of IA  42  should be exactly zero; however, due to previously described effects, the voltage at the input terminals of IA  42 , after passage through the sense connections and RFI filter  38 , contains some systematic offset error (mostly due to thermoelectric voltages generated at junctions of dissimilar materials having different temperatures). At point  82  the RFI filter  38  starts to settle toward the final stable value. 
     At point  83  the RFI filter  38  has finished settling and provides a stable value to the IA  42 ; the output voltage of IA  42  is sampled by one of the channels of A/D  47  and/or one of the D/A  46  outputs is changed until output from IA  42  is equal to zero. 
     At point  84  the calibration for this channel  74 / 38 / 40 / 42  is finished. 
     At points  85  the MOSFET switch  74  is opened. 
     At point  86  the RFI filter  38  has fully settled to the value representative of the current flowing in shunt  72 . 
     Finally at  87  the output is switched to the newly calibrated channel  74 / 38 / 40 / 42  by changing the state of signal Mux  48 ; now the output terminals  44  are connected to the output of IA  42 . 
     Note that for the duration of the calibration of the measurement channel related to closed switch  74 , the channel related to open switch  73  supplied the output signal. 
     At points  87  through  93  exactly the same procedure as described above, from point  81  to point  87 , is applied to the measurement channel  73 / 37 / 39 / 41 , with actions applied to MOSFET  73  and Mux  48 . 
     Finally, a new algorithmic cycle starts at point  93 . 
     It will be appreciated that at the very beginning of the operations (when the power is first applied to the whole circuit), a slightly different cadence could be utilized in order to acquire the initial values of the offset errors; then the complete cycle  80  repeats indefinitely. However the duration of the complete calibrate/measure/settle/switchover cycle  80  can be adjusted, depending on the rate of change of the temperature and/or temperature gradients over the circuit. If it is desired to adjust this cycle, for example to reduce the amount of time spent in calibration and zeroing activity, then the temperature should be sensed with a suitable sensor or sensors. Such sensors are omitted for clarity in  FIG. 3 . 
     It will likewise be appreciated that if the amount of time spent in calibration and zeroing is reduced, this makes it possible for some periods of time when both signal processing channels are in measurement service. This permits cross-correlating the two measured signals (at A/D  47 ), whether for purposes of averaging or for purposes of early warning of possible failure in one of the two channels. 
     In addition, temperature measurements will be used to compensate for the temperature dependency of the resistance of the current shunt  72 . For example, in a physical embodiment as shown in  FIG. 5 , the current shunt is created with a copper pattern on a PCB (and thus avoids the necessity to procure and assemble separate discrete components); in this case the dependency of the copper used for the conductive tracks on the PCB would be well known and predictable, and the resistance value changes could be readily compensated by digital calculations in the control circuit  51 . If required, the continuous analog output at output terminals  44  can also be adjusted to compensate for the current shunt&#39;s resistance changes; however, the main application of the continuous analog output is to provide the fast electronic solid state fuse capability which may not require very high precision, and may well be served sufficiently with a value that has not been compensated for resistance changes with temperature. Nevertheless, the circuit required for the adjustment of the continuous analog output will involve utilization of programmable-gain functionality in IA  41  or  42 ; this will be familiar to the reader. 
     Likewise, the initial full-scale error created by an inaccurate value (due to manufacturing tolerances) of the shunt  72  at room temperature, may be calibrated out by passing an accurately known current through the shunt  72 , and recording (preferably in the local non-volatile memory) an appropriate value for the digital adjustment of gain for the whole current sensor system. 
     The current invention provides sufficient and arbitrarily long time for settling of the RFI filters  37  and  38 , and is able to accommodate filters with extremely long time constants. This is beneficial for the accuracy of the measurements when output of the current sensor is used for accumulation of the total charge passed through the input terminals  30  (e.g. Coulomb counting). If such an integrated current value is maintained and calculated in the digital part of the circuit, an increase in the time constant of the RFI filter can be taken advantage of for lowering of the required sampling rate of the A/D converters  47  ( FIG. 3 ). 
     However, in this case, the fast response time of the (analog) output, used for solid-state fuse functionality, will be compromised. 
     A solution for this problem according to the current invention is disclosed in  FIG. 6 . This drawing shows one channel of measurements that contains two RFI filters  37  and  52 , and two corresponding amplifiers  41  and  54 ; all of these are associated with switch  73 . The other channel (related to switch  74 ) is not shown for clarity of the drawing; nonetheless, it also contains two RFI filters and two instrumentation amplifiers. The output of the IA  41  is connected to the output terminals  44  via Switch  43  (as shown in  FIG. 3 ); RFI filter  37  has a time constant that is appropriate for the solid-state fuse functionality. On the other hand, the output of the IA  54  is only connected to the additional A/D input at the control circuit  51 , the value on the output of IA  54  will only be reported in digital form after processing in the control circuit  51 , and it does not require the offset-compensation D/A (as the calibration value would be subtracted digitally). In this signal processing channel, the RFI filter  52  has a time constant that is suitable for the measurements of the charge passing through the current shunt  72 , while utilizing a slow sampling rate of the A/D. RFI filter  52  has a time constant that is many times longer than the time constant of RFI filter  37 . The timing of the operating algorithm (as illustrated in  FIG. 4 ) is thus set to properly service the timing requirements of the RFI filter  52  (since RFI filter  37  has a shorter time constant, its settling time requirements will certainly be satisfied). 
     As mentioned above, in this embodiment exactly the same circuit is constructed after switch  74 , with two RFI filters and two instrumentation amplifiers. An IA that is associated with a fast RFI filter can be connected to output terminals  44  via switch  43 , while the output of other IA is connected only to an additional A/D input of the control circuit  51 . 
     Accordingly, the embodiment disclosed in  FIG. 6  can be used simultaneously for fast electronic solid state fuse functionality via a continuous analog output, and for precise measurements of the accumulated charge that is reported digitally by control circuit  51 . 
     It might be asked whether the same functionality (serving both the fast-fuse need and the need for very accurate measurement) could be achieved by providing a fast RFI filter, followed by an IA, and then followed by another filter that is suitable for reduction of the A/D sampling rate, as is typical of older circuits. It turns out that this does not work well, and that it really is better to provide the fast and slow signal processing paths in parallel as shown in  FIG. 6 . To understand why this does not work well, consider that if the current is to be measured accurately, the fast transients of the measured current would need to be faithfully reproduced and amplified by the first  1 A, and from this amplified result, the bandwidth of the signal would need to be reduced by the second (slow) filter. But if there is any nonlinearity in that first instrumentation amplifier, with such nonlinearity exacerbated by possibly quickly changing signals, then this could create clipping and could generally disturb the otherwise intended linear operations of the IA. Such clipping and nonlinearity would have a profound effect on the accuracy of the measurements. 
     By comparison, the approach shown in  FIG. 6  has a “slow” signal processing path that maintains accuracy and is not dependent upon any requirement of high performance by the “fast” signal processing path. In the approach shown in  FIG. 6 , so far as the accurate but slow path is concerned, the circuit firstly limits the bandwidth with a “slow” filter  52  (utilizing simple passive components for the filter  52  that are naturally highly linear even in the presence of transients). Then the output of the “slow” filter  52  is amplified by IA  54 . It is noted that IA  54  can thus be a relatively low-bandwidth unit, with correspondingly very low power consumption. 
     It will be recognized that the control circuit  51  will be best served by the use of a microprocessor; the whole circuit  51  can in fact be contained in a single-chip microcontroller that also includes the required functionality of A/D  47 , D/A  46 , communication interface  50 , and port pins that will serve as Mux signal  48  and FET Drivers  45 . The instrumentation amplifiers can reside on the same chip for a compact and inexpensive solution. 
     Alternate circuits are shown in  FIG. 7  and  FIG. 8 . These circuits replace the resistors  75   a - 75   d  of  FIG. 3 ,  FIG. 5  and  FIG. 6  with FETs  96 / 97 . Depending on exact implementation details and requirements, the circuits in  FIG. 7  and  FIG. 8  may have advantages over the circuits in  FIG. 3  and  FIG. 6 , and vice versa. 
     The alert reader will have no difficulty devising myriad obvious variations and improvements upon the invention, after having appreciated the teachings herein. It is intended that the claims which follow will embrace such variations and improvements. 
     The alert reader will appreciate that what has been described is at least two ways in which the first and second measurement points could be connected with the first and second input terminals of the first instrumentation amplifier, which ways could be termed an impedance means. One way that the impedance means could be accomplished is by arranging for impedance means to comprise a first resistor connecting the first measurement point with the first input terminal of the first instrumentation amplifier and a second resistor connecting the second measurement point with the second input terminal of the first instrumentation amplifier. Another way that the impedance means could be accomplished is by arranging for impedance means to comprise a third switch connecting the first measurement point with the first input terminal of the first instrumentation amplifier and an electrical connection connecting the second measurement point with the second input terminal of the first instrumentation amplifier. 
     The same could be said for a second impedance means connecting the first and second measurement points to the first and second input terminals of the second instrumentation amplifier. Such an impedance means could for example be accomplished by such resistors or by such switches and electrical connections.