Abstract:
An electronic device has an LDO regulator for varying loads. The LDO regulator includes a primary supply node coupled to a primary voltage supply. An output node provides a secondary supply voltage and a load current. A bias current source generates a bias current. A gain stage coupled to the bias current source increases the maximum available load current. The gain stage includes a first MOS transistor biased in weak inversion coupled to a current mirror which mirrors the drain current through the first MOS transistor to the output node. The gate-source voltage of the first MOS transistor increases in response to a decreasing secondary supply voltage level at the output node to increase the available load current.

Description:
CLAIM OF PRIORITY 
   This application claims priority under 35 U.S.C. 119(a) to German Patent Application No. 10 2007 0041 155.5 filed Aug. 30, 2007 and 35 U.S.C. 119(e)(1) to U.S. Provisional Application No. 61/016,890 filed Dec. 27, 2007. 

   TECHNICAL FIELD OF THE INVENTION 
   The technical field of this invention is an LDO for use in an electronic device and more particularly, an LDO regulator with a high dynamic range for varying loads. 
   BACKGROUND OF THE INVENTION 
   A key parameter for microcontroller based applications and almost all applications including portable or mobile electronic devices is the current consumption in a low power mode (LPM). While the electronic system is in a low power mode, the CPU is typically idle and does not execute a program. The system consumes only an absolute minimum of current, just as much as is necessary in order to keep the system operable. Some applications need low drop out voltage regulators (LDOs) providing regulated supply voltages. The regulated supply voltage provided by the LDO must be maintained even during a LPM phase. Since supply current is limited and is the most valuable resource in the system, the current consumed by the LDO must be extremely low during LPM phases. In LPM phases the LDO is expected to consume and provide currents which are only in the order of nano Amperes (nA). However, there might be special situations, even in LPM where the LDO must provide currents that can be orders of magnitudes greater, for example several tens of micro Amperes (μA). 
   BACKGROUND OF THE INVENTION 
   This invention is an electronic device with an LDO which provides a large dynamic range of the load current while having very low self power consumption. 
   The present invention is an electronic device having an LDO regulator for varying loads. The LDO regulator has a primary supply node coupled to a primary voltage supply and an output node providing a secondary supply voltage and a load current. A bias current source generates a bias current. A gain stage coupled to the bias current source increases the maximum available load current. The gain stage includes a first MOS transistor biased in weak inversion. This first MOS transistor is coupled to a current mirror mirroring the drain current through the first MOS transistor to an output node. Further, the gate source voltage of the first MOS transistor increases in response to a decreasing secondary supply voltage level at the output node to increase the available load current. The bias current generated by the bias current source drives the first MOS transistor. The drain current of the first MOS transistor is mirrored using the current mirror so that the current received at the output node is proportional to the bias current. When the voltage at the output node (the secondary supply voltage) decreases the gate source voltage of the first MOS transistor increases because the first MOS transistor is biased in weak inversion (i.e. the gate voltage applied to the first MOS transistor is less than its threshold voltage). The current mirrored from the first MOS transistor to the output node increases, which increases the size of the load current at the output node. In this way, the LDO regulator only needs a very low current (e.g. about 100 nA to 300 nA) for its own operation and yet is able to drive a current of several tens of μA (for example 30 μA) as load current when in low power mode (LPM). The present invention thus allows the lowest supply current to be used, but is also able to deliver load currents that are orders of magnitude higher than in the unloaded case. 
   Preferably, the first MOS transistor has a gate coupled to a constant reference voltage level and a source coupled to a first node. The voltage level of the first node drops in response to the decreasing secondary supply voltage level at the output node. Thus the secondary supply voltage level at the output node is fed back to the gain stage. This causes the voltage at the first node to decrease when the voltage level at the output node decreases. This causes the gate source voltage of the first MOS transistor to increase further. 
   The gain stage may include a second MOS transistor and a third MOS transistor. The gate of the second MOS transistor is coupled to the output node, with a source of the second MOS transistor and a drain of the third MOS transistor coupled to the first node. A drain of the second MOS transistor is coupled to the bias current source and a gate of the third MOS transistor is coupled to the drain of the second MOS transistor. The secondary supply voltage at the output node is then the voltage applied to the gate of the second MOS transistor. Thus, as the secondary supply voltage decreases, the gate voltage of the second MOS transistor decreases and the amount of current from the bias current source through the second MOS transistor decreases, leading to a voltage decrease at the first node. 
   The current mirror preferably comprises a diode connected fourth MOS transistor and a fifth MOS transistor having a gate coupled to a gate of the fourth MOS transistor and biased in weak inversion. A drain of the fourth MOS transistor is coupled to a drain of the first MOS transistor and a source of the fourth MOS transistor coupled to a resistive element such that the gate source voltage of the fifth MOS transistor corresponds to combined voltages of both the gate source voltage of the fourth MOS transistor and a voltage drop across the resistive element. The fourth and fifth MOS transistors then form the current mirror and mirror the current from the first MOS transistor to the output node. 
   In another aspect of the present invention includes a sixth MOS transistor. The gate of the third MOS transistor is coupled through the sixth MOS transistor to the drain of the third MOS transistor. A drain of the sixth MOS transistor is coupled to the gate of the third MOS transistor. A source of the sixth MOS transistor is coupled to the drain of the second MOS transistor. The source of the sixth MOS transistor is further coupled to a second bias current source. A gate of the sixth MOS transistor receives a constant voltage level. The sixth MOS transistor closes the feedback loop to the third MOS transistor without restrictions on the voltage input range and has a common gate configuration so that the dominant pole of the feedback loop will be at the gate of the third MOS transistor. The stability of the LDO circuit is then assured since all circuit loops are single pole only. Addition of the sixth MOS transistor to the feedback loop increases the voltage input range to the gain stage fed back from the output node. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and other aspects of this invention are illustrated in the drawings, in which: 
       FIG. 1  is a simplified circuit diagram of an LDO regulator according to a first embodiment of the invention; 
       FIG. 2  is a simplified circuit diagram of an LDO regulator according to a second embodiment of the invention; 
       FIG. 3  is a logarithmic graph of supply current as a function of load current for an LDO regulator according to the invention; and 
       FIG. 4  is a logarithmic graph of LDO output voltage as a function of load current for an LDO regulator according to the invention. 
   

   DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     FIG. 1  shows a simplified circuit diagram of an LDO regulator according to a first embodiment of the invention. The LDO regulator shown is for use in an electronic device such as a microcontroller. 
   Primary supply voltage node AVDD is connected to a primary voltage supply, the DC voltage supply of the device including the LDO regulator. Supply voltage node AVDD is connected to bias current generator I B1 , which generates a bias current I BIAS , resistor R 0  and the source terminal of PMOS transistor MP 5 . Resistor R 0  is connected to the source terminal of another PMOS transistor MP 4 . The gate terminals of transistors MP 4  and MP 5  are interconnected so that transistors MP 4  and MP 5  form a current mirror stage. Transistor MP 4  is diode connected; i.e., its gate and drain terminals are interconnected. Both bias current generator I B1  and the current mirror stage are connected to a gain stage GS. Gain stage GS includes first, second and third NMOS transistors MN 1 , MN 2  and MN 3 . First NMOS transistor MN 1  has a drain terminal connected with the gate and drain of transistor MP 4  in the current mirror stage. The gate of first NMOS transistor MN 1  is connected to reference voltage source V REF . The source terminal of transistor MN 1  is connected to the source terminal of second NMOS transistor MN 2  and to the drain terminal of third NMOS transistor MN 3  at node K 1 . The source terminal of transistor MN 3  is connected to ground. The gate terminal of transistor MN 3  is connected to a node interconnecting bias current generator I B1  and the drain terminal of transistor MN 2 . The drain terminal of transistor MP 5  at the output of the current mirror stage is connected to an output node V OUT , which provides a secondary supply voltage and a load current (I LOAD ). The current mirror stage formed of transistors MP 4  and MP 5  mirrors current from transistor MN 1  in the gain stage GS to output node V OUT . Output node V OUT  is also connected to the gate terminal of transistor MN 2  forming a feedback loop to gain stage GS. Load capacitor C LOAD  is connected between output node V OUT  and ground. 
   Initially, load current I LOAD  at output node V OUT  is low and is of the order of current I BIAS  generated by bias current source I B1 . Transistor MN 2  is driven by bias current I BIAS . Due to the gate voltages of transistors MN 1  and MN 2  being about the same (the gate voltage of transistor MN 1  is reference voltage V REF ), a current I BIAS  also flows through transistor MN 1  if transistors MN 1  and MN 2  are symmetrical. The current through transistor MN 1  is mirrored by the current mirror stage MP 4 , MP 5  and R 0  to output node V OUT . The output voltage at output node V OUT  is fed back to the gain stage GS at the gate of transistor MN 2 . The drain current through transistor MN 3  is controlled by a regulation loop provided by the gate of transistor MN 3  being connected to the bias current source I B1  and can be chosen equal to twice the bias current I BIAS . Since the output is initially loaded only with a very small load current, which is about equal to the bias current I BIAS , the gate-source voltage of transistor MP 5  in the current mirror stage is approximately equal to the gate source voltage of transistor MP 4  in the current mirror since the voltage drop across the resistor R 0  can be neglected for small currents. Thus:
 
 V   GS   *MP 5 =V   GS   *MP 4.
 
As load current I LOAD  at output node V OUT  becomes larger, the output voltage, or secondary supply voltage at the output node V OUT  will eventually decrease. The decrease in output voltage fed back to the gate of transistor MN 2  therefore causes the node K 1  to be pushed to lower voltages. This opens the gate source voltage of transistor MN 1 . Thus the gate source voltage of transistor MN 1  and therefore the current flowing through transistor MN 1  will increase. This means that the gate source voltage of transistor MP 5  in the current mirror will become equal to the gate source voltage of transistor MP 4  plus the voltage across the resistor R 0 . This boosts the current through transistor MP 5 :
 
 V   GS   *MP 5 =V   GS   *MP 4 +V   R0 .
 
The sum of the currents flowing through transistors MN 1  and MN 2  will then be received at transistor MN 3 . This is controlled by the regulation loop. In other words, the decrease in output voltage at output node V OUT  increases the gate source voltage at transistor MN 1 , and therefore at transistor MP 5  in the current mirror. These transistors MN 1  and MP 5  are in deep subthreshold, because of being biased in weak inversion. When their gate source voltages are changed there will be an exponential increase of drain currents in both transistors MN 1  and MP 5 . Therefore this circuit offers a large dynamic range of output currents at the drain of transistor MP 5  and thus at the output node V OUT  for just a small drop of output voltage at output node V OUT .
 
   Without an external load current, the LDO circuit operates with a very low bias current I BIAS  of the order of 10 nA. Overall the LDO consumes a supply current I SUPPLY  of between 200 nA and 300 nA. In terms of external current loading, the LDO can deliver a load current I LOAD  that is orders of magnitude higher than the bias current I BIAS . Therefore the LDO achieves both a low current consumption at a low I SUPPLY  and a high potential load current drive in combination. 
   In  FIG. 1 , the other feedback loop controlling the gate voltage of transistor MN 3  is directly connected to the drain of transistor MN 2 . This means that the voltage input range at the gate of transistor MN 2  is limited due to the feedback connection of transistor MN 3 .  FIG. 2  shows a second embodiment of the invention that overcomes this drawback of the circuit in  FIG. 1 . The LDO circuit shown in  FIG. 2  is almost the same as that shown in  FIG. 1 , except that the bias current source I B1  is moved from the position shown in  FIG. 1 , between the supply voltage node AVDD and the drain of transistor MN 2 , and is instead connected between the gate of transistor MN 3  and ground. A second current source I 2  is then connected between the supply voltage node AVDD and the drain of transistor MN 2  in place of the bias current source I B1 . A node interconnecting the gate of transistor MN 3  and the bias current generator I B1  is connected to the drain of an additional PMOS transistor MP 6 . The source of transistor MP 6  is connected to a node interconnecting the current source I 2  and the drain of transistor MN 2 , with the gate of transistor MP 6  being connected to a constant voltage source V 1 . 
   The additional transistor MP 6  closes the feedback loop to transistor MN 3  without the restrictions on the voltage input range exhibited by the LDO circuit of  FIG. 1 . Since transistor MP 6  is in a common gate configuration, the dominant pole of the loop will be at the gate of transistor MN 3 . There will always be sufficient phase margin and the stability of this circuit is always assured, since both of the feedback loops V OUT -MN 2 -MN 1 -MP 4 -MP 5  and MN 3 -MN 2 -MP 6  only have a single pole. The outer feedback loop from the output voltage node V OUT  (V OUT -MN 2 -MN 1 -MP 4 -MP 5 ) is dominated by the load capacitor C LOAD . Load capacitor C LOAD  preferably has a capacitance of 470 nF in this example. The inner loop (MN 3 -MN 2 -MP 6 ) has one pole at the gate of transistor MN 3 . 
     FIGS. 3 and 4  show the DC response of the LDO circuit for the circuit shown in  FIG. 2 . The circuit shown in  FIG. 1  has basically the same behavior.  FIG. 3  illustrates load current I LOAD  in terms of supply current I SUPPLY  on a logarithmic scale.  FIG. 4  illustrates load current I LOAD  in terms of the output voltage at the output voltage node V OUT  on a semi-logarithmic scale. In this example, reference voltage V REF  applied to the gate terminal of transistor MN 1  is 1.8 V. When the load current I LOAD  at the output voltage node V OUT  is near or equal to zero, the supply current is around 300 nA. As the load current I LOAD  increases, the LDO output voltage V OUT  decreases and it can be seen that the circuit can deliver a load current I LOAD  of up to about 100 μA. 
   Although the present invention has been described with reference to specific embodiments, it is not limited to these embodiments and no doubt further alternatives will occur to the skilled person that lie within the scope of the invention as claimed.