Abstract:
A buffer employs an input stage with an active, LC load. The active load includes integrated inductors that combine with the parasitic gate capacitances of a pair of transistors in a negative-transconductance (−Gm) booster configuration. The resulting active load emphasizes a desired frequency, improving the quality, or “Q,” of the input stage, and consequently of the entire buffer.

Description:
BACKGROUND 
     Modern high-speed digital communication systems employ data transmit and recovery circuits operating at or above ten gigabits per second. Due to the bandwidth limitations of conventional CMOS processes, it is difficult to produce an amplifier capable of driving a significant load with better than unity gain at frequencies above about five gigahertz (GHz). At frequencies approaching five GHz, amplifier gain is at the tail portion of the roll-off characteristic. 
     FIG. 1A (prior art) is a Bode plot depicting a roll-off characteristic for a typical CMOS amplifier. The roll-off (−3 dB) frequency F o  is around several hundred MHz. More exotic processes, such as those employing silicon germanium or gallium arsenide, provide improved high-frequency response; unfortunately, this improvement comes at considerable expense. 
     FIG. 1B (prior art) depicts a communication system  100  that includes a transmitting integrated circuit (IC)  105  and a receiving IC  110 . Transmitting IC  105  includes an output amplifier  115  driving an external pin  120  via a bond pad  125  and a bond wire  130 . An electrostatic-discharge (ESD) circuit  135  connects to bond pad  125  and the output of amplifier  115  to protect IC  105  from damage due to ESD events. IC  110  includes an input amplifier  140  that receives the output of IC  105  via a printed-circuit-board (PCB) trace  145 , an input pin  150 , a bond wire  155 , and a bond pad  160 . IC  110  also includes an ESD circuit  165  connected to the signal line between pad  160  and the input to amplifier  140 . 
     System  100  illustrates that amplifiers intended to drive signals off chip must contend with capacitive loading far greater than normally experienced on chip. Such signals encounter capacitive loading from e.g. bond pads  125  and  160 , bond wires  130  and  155 , ESD circuits  135  and  165 , PCB trace  145 , and the input of amplifier  140 . These capacitances collectively shift the pole of output amplifier  115  toward zero, exacerbating the problem of communicating at high frequency. 
     Also problematic, increased load capacitance reduces amplifier bandwidth. The unity-gain bandwidth of amplifier  115  is defined by g m /C ld , where g m  is the amplifier transconductance and C ld  is the capacitive load on the amplifier. Load capacitance C ld  is typically in the neighborhood of 1.2 pf, providing a bandwidth typically in the range of several hundred megahertz. 
     An article by Savoj and Razavi entitled “A 10 Gb/s CMOS Clock and Data Recovery Circuit with Frequency Detection,” 2001 IEEE International Solid-State Circuits Conference, describes a CDR circuit that addresses the problem of providing high-frequency signals off chip using relatively inexpensive CMOS processes. That article is incorporated herein by reference. 
     FIG. 2 (prior art) depicts the output buffer (amplifier)  200  described in the Savoj and Razavi article. Buffer  200  includes an input stage  205  and an output stage  210 . Output stage  210  experiences the capacitive loading described above in connection with FIG.  1 B. The transistors associated with output stage  210  are relatively large, helping output stage  210  contend with the load. Due to their size, the transistors of output stage  210  present a significant capacitive load (e.g., 0.5 picofarads) to input stage  205 . Input stage  205  employs inductive peaking to increase high-frequency, small-signal gain in the face of the input capacitance of output stage  210 Still, there is always a demand for higher performance. 
     SUMMARY 
     A buffer in accordance with the invention employs an input stage with an active, LC load. The active load includes integrated inductors that combine with the parasitic gate capacitances of a pair of transistors in a negative-transconductance (−Gm) booster configuration. The resulting active load emphasizes a desired frequency, improving the quality, or “Q,” of the input stage, and consequently of the entire buffer. 
     This summary does not define the scope of the invention, which is instead defined by the appended claims. 
    
    
     BRIEF DESCRIPTION OF THE FIGURES 
     FIG. 1A (prior art) is a Bode plot depicting a roll-off characteristic for a typical amplifier. 
     FIG. 1B (prior art) depicts a communication system  100  that includes a transmitting IC  105  and a receiving IC  110 . 
     FIG. 2 (prior art) depicts a conventional high-speed output buffer  200 . 
     FIG. 3A depicts a buffer  300  in accordance with an embodiment of the invention. 
     FIG. 3B is a Bode plot  380  illustrating the hypothetical frequency response of a buffer like buffer  300  of FIG.  3 A. 
     FIG. 4A depicts a buffer  400  adapted in accordance with another embodiment of the invention. 
     FIG. 4B depicts a resistor calibration circuit  425  used to control the resistance through each leg of the input stage of buffer  400  of FIG.  4 A. 
     FIG. 5 schematically depicts a resistor calibration circuit  500  in accordance with another embodiment of the invention. 
    
    
     DETAILED DESCRIPTION 
     FIG. 3A depicts a buffer  300  in accordance with an embodiment of the invention. Buffer  300  includes a pair of differential input transistors  305  and  310 , the sources of which are connected to a power-supply terminal (ground) via a current source  315 . The drains of transistors  305  and  310  connect to another power-supply. terminal (VDD) via a load that includes three resistors  320 - 322 , a pair of inductors  325  and  330 , and a pair of cross-coupled PMOS transistors  335  and  340 . Buffer  300  includes an output stage with a pair of differential NMOS transistors  345  and  350 , the sources of which are connected to ground via a current source  355 . The drains of transistors  345  and  350  are connected to respective buffer output terminals OUT_P and OUT_N, and to VDD via respective termination resistors  360  and  365 . 
     Resistor  322  provides the appropriate bias voltage for transistors  335  and  340 . Parasitic gate capacitances  370  and  375  are connected in parallel, each having a first terminal connected to the upper terminal of resistor  320  and a second terminal connected to the upper terminal of resistor  321 . Inductors  325  and  330  are connected in series between the same two terminals, so that inductors  325  and  330  combine with capacitances  370  and  375  to form an LC circuit in which the resonant frequency F R  is about:                F   R     =     1     2                 π        LC                 Eq.   1                                
     where L is the inductance of each of inductors  325  and  330  and C is the value of each gate capacitance  370  and  375 . Other circuit features, such as resistors  320  and  321  and the gate capacitances associated with transistors  345  and  350 , also play a role in determining the resonant frequency; however, these effects are ignored here for simplicity. 
     Transistors  335  and  340  act as a negative-Gm (−Gm) amplifier, amplifying the signals presented on their respective gates. This amplification boosts the Q of the above-described LC circuit, improving the gain response of buffer  300 . 
     FIG. 3B is a Bode plot  380  illustrating the hypothetical frequency response of a buffer like buffer  300  of FIG.  3 A. Bode plot  380  includes a plot  385  of amplifier  300  without an inductive LC load and a plot  390  of the frequency response of the LC filter formed principally by inductors  325  and  330  and capacitances  370  and  375 . The overall response (plot  395 ) of buffer  300  is a combination of the responses depicted in plots  385  and  390 . 
     In a specific embodiment fabricated using a 0.18-micron process, resistor  322  is 26 ohms, resistors  360  and  365  are external fifty-ohm termination resistors, and resistors  320  and  321  are integrated 70-ohm resistors. Current source  355  can be impedance matched to resistors  360  and  365 . 
     Inductors  325  and  330  are fabricated as a single, center-taped, 3-turn inductor using the top metal layer. The shape of the inductor is similar to those shown in FIG. 5.3.7 of the above-referenced article by Savoj and Razavi: the inductor is a square “coil” in which the wire width is about 26 microns spaced about 3 microns from adjacent turns of the coil. The resulting inductance is about 2.7 nH. Together with capacitances  335  and  340 , inductors  325  and  330  provide a resonant frequency of about 5 GHz. Simulation suggests this embodiment can drive a 1.3 pF load across terminals OUT_P and OUT_N, with worst-case 600 mV swing, single ended, at 10 Gb/s. 
     Process variations can significantly affect the values of resistors  320  and  321 , and consequently the DC gain of buffer  300 . FIGS. 4A and 4B depicts a buffer  400  adapted in accordance with another embodiment of the invention to address this problem. Buffer  400  is similar to buffer  300  of FIG. 3A, like-numbered elements being the same. Buffer  400  differs from buffer  300  in that resistors  320  and  321  are replaced with a pair of adjustable, calibrated resistors, each of which includes a resistor  405  connected in parallel with a transistor  410 . The gates of transistors  410  are both connected to a control voltage VB that establishes the appropriate resistance through transistors  410 , and consequently maintains the appropriate DC gain for the input stage of buffer  400 . 
     Buffer  400  also includes a pair of resistors  415  and  420 , 10,000 ohms each in one embodiment. Resistors  415  and  420  define between them a common-mode reference node RB that, with reference node RA between inductors  325  and  330 , is used to derive a feedback signal for controlling the resistances through transistors  410 . 
     FIG. 4B depicts a resistor calibration circuit  425  used to develop control voltage VB, and consequently to control the resistance through each leg of the input stage of buffer  400  of FIG.  4 A. Calibration circuit  425 , transistors  410 , and resistors  405  provide calibrated resistances between the nodes on either side or resistors  405 . Calibration circuit  425  includes three differential amplifiers  427 ,  429 , and  431 ; three NMOS transistors  433 ,  435 , and  437 ; a PMOS transistor  439 ; and a pair of resistors  441  and  443 . Resistor  443  is defined in the fabrication sequence to be identical to resistors  405 ; likewise, transistor  433  is defined in the fabrication sequence to be identical to transistors  410 . Resistor  441  is a precision resistor, in one embodiment a 60-ohm external resistor. As explained below, resistor  441  acts as a reference to establish the resistances of transistors  410 , and consequently the DC gain of the input stage of buffer  400 . 
     Amplifiers  427  and  429  have their non-inverting input terminals connected to respective reference nodes RA and RB of buffer  400 . Amplifier  427  and PMOS transistor  439  together form a unity-game amplifier that provides the voltage on reference node RA to the top terminals of resistors  441  in  443 , and to the drain of transistor  433 . Amplifier  429  and NMOS transistor  437  form another unity-game amplifier that provides the voltage on reference node RB to the bottom terminal of resistor  443 , the source of transistor  433 , and the non-inverting input terminal of amplifier  431 . 
     Transistor  435  mirrors the current through transistor  437 , so the current through reference resistor  441  equals the sum of the currents through resistor  443  and transistor  433 . Amplifier  431  controls the gate of transistor  433  so that both the inverting and non-inverting inputs to amplifier  431  are at the same potential, so the same voltage is applied across resistor  441 , resistor  443 , and transistor  433 . The combined resistance through resistor  443  and transistor  443  is therefore controlled to be equal to the reference resistance of resistor  441 . The control voltage VB provided to the gate of transistor  443  to establish this equivalent resistance is also provided to the gates of transistors  410  to control the resistance through each leg of the input stage of buffer  400 , and therefore the gain of proper  400 . 
     FIG. 5 schematically depicts a resistor calibration circuit  500  in accordance with another embodiment of the invention. Calibration circuit  500  includes five differential amplifiers  505 ,  510 ,  515 ,  520 , and  525 ; two PMOS transistors  530  and  535 ; five NMOS transistors  540 ,  545 ,  550 ,  555 , and  560 ; and four resistors  565 ,  570 ,  575 , and  580 . Each of amplifiers  505 ,  510 ,  515 , and  520  connects to a respective transistor in a unity-game configuration. The voltages on the top and bottom terminals of resistors  565  in  570  are therefore the voltages on reference nodes RA and RB, respectively, of buffer  400  of FIG.  4 A. 
     Resistor  565  is an external, precision resistor. The current through resistor  565  is mirrored through resistor  575  by transistors  540  and  550 . Similarly, the combined current through resistor  570  and transistor  560  is mirrored through resistor  580  by transistors  545  in  555 . Resistors  575  and  580  are matched, so the voltages on the input terminals of amplifier  525  are equal when the current through resistors  575  and  580  are equal. Amplifier  525  produces the appropriate control voltage VB on the gate of transistor  560  to equalize the resistances through resistor  565  and through the combined resistor  570  and transistor  560 , and consequently the voltages on the input terminals to amplifier  525 . The control voltage VB required to establish this equivalence is conveyed to the gates of transistors  410  of buffer  400  (FIG. 4A) to control the gain of the input stage of buffer  400 . 
     While the present invention has been described in connection with specific embodiments, variations of these embodiments will be obvious to those of ordinary skill in the art. Moreover, some components are shown directly connected to one another while others are shown connected via intermediate components. In each instance, the method of interconnection establishes some desired electrical communication between two or more circuit nodes, or terminals. Such communication may often be accomplished using a number of circuit configurations, as will be understood by those of skill in the art. Therefore, the spirit and scope of the appended claims should not be limited to the foregoing description.