Abstract:
An arrangement of hybrid couplers, selectively removing RF amplifiers from a receiving path from an antenna back to a receiver in response to an event, the event switching impedances of switchable impedances connecting to ports of the hybrid couplers, between a matching impedance and mismatching impedance. The matching and mismatching impedances change coupling characteristics of the hybrid couplers between a normal coupling and a bypass coupling. In the normal coupling the hybrid couplers provide port-to-port coupling placing the RF amplifiers within the receiving path and, in the bypass coupling, the hybrid couplers provide port-to-port coupling bypassing the RF amplifiers. Optionally, transmitted signal power is detected to identify transmit and receiving intervals. During detected transmit intervals, isolation switches isolate the RF amplifiers from the feedback path and, during receiving intervals and, during detected receiving intervals, the isolation switches RF amplifiers into the feedback path to amplify reception signals. Optionally a hybrid coupler switch is placed between the two circulators to increase the isolation in the RX mode reducing feedback to the LNA&#39;s.

Description:
TECHNICAL FIELD 
     Embodiments relate generally to time division duplex connections of radio frequency transmitters, receivers, related amplifiers, and antenna elements. 
     BACKGROUND 
     Wireless radio frequency (RF) communications may employ a time division sharing or multiplexing of a common antenna by RF transmitting units and RF receiving units. For purposes of this description, the term “antenna,” standing alone, means at least major radiating element(s) of what may be a larger structure. Such time division sharing may be employed by, for example, the RF receiving and RF transmitting units forming the transceiver systems within a base station of a wireless communication system, e.g., a cellular telephone system. Because the sharing of a common antenna is between signals traveling in opposite sending and receiving directions, this may be referred to as Time Division Duplex (TDD) sharing. 
     To achieve a desirable coverage area, antenna are often positioned a considerable height above ground. The positioning may be accomplished by, for example, mounting the antenna on a tower. In such a mounting arrangement, the RF receiving unit may not be positioned proximal to the antenna feed. Instead, it may be located at a distance away that is at least equal to the height of the antenna. However, as known in the RF communication arts, propagating an un-amplified reception signal from an antenna feed to a receiver located a significant distance away may results in a signal, when arriving at the receiver, having substantially lower signal-to-noise ratio (SNR) that exhibited at the feed. To avoid such degradation, an RF amplifier may be arranged between the feed port of the antenna and the input of the RF receiving unit, configured to amplify the RF signal received from the antenna feed port into a higher power signal, thereby reducing the degradation in signal-to-noise ratio (SNR) caused by additive noise. Typically, for reasons well known to persons skilled in the relevant arts, the RF amplifier that amplifies the RF signal received from the antenna feed port for transmission to the RF receiving unit is a low noise amplifier (LNA). 
     The receiving RF LNA, however, often imposes a requirement that signal energy from the RF transmitter does not couple onto input of the RE LNA during operation of the RF receiver unit. Such energy would be amplified by the RF amplifier and then input to the RE receiver unit. 
     One known arrangement for reducing RF transmitter energy coupling into the receiving RF LNA input is a switching circuit that alternately connects the output of the transmitting unit and the input of the RF LNA amplifier to the feed terminal of the antenna. For brevity and consistency, this prior art switching arrangement will be arbitrarily labeled as a “switch-based TDD RF LNA isolator.” 
     Prior Art  FIG. 1  shows an exemplar of a switch-based TDD RE LNA isolator, connecting between a base station transceiver (BST) port  10  and an antenna feed port  16 . The feed to the BST port  10  is not shown. For example, BST port  10  may connect to the input (not shown) of a receiver unit (not shown) and to the output (no shown) of a transmitter unit (not shown). In the transmit mode the switches  20  and  24  isolate the RF amplifier  18  and connect the BST port  10  to the antenna port  16 . In contrast, in the receive mode, the switches  20  and  24  isolate the BST port  12  from the antenna port  16 , while providing a path for incoming RF communication signals from the antenna feed port  16  to the input  18 A of the RF amplifier  18 , and a path from the output  18 B of the RF amplifier  18  to the BST port  10 . 
     Another known apparatus and method for isolating signal output of the RF transmitter from the input of the RF receiving amplifier employs a particular arrangement of circulators. The operation of circulators is known to persons skilled in the related arts and, therefore, a detailed description is omitted here. Prior Art  FIG. 2  illustrates an exemplar of a known circulator-based TDD RF LNA isolating apparatus. Referring to  FIG. 2 , the illustrated exemplar has a first circulator  12  fed at one port ( 12 A) by the BST port  10 , and a second circulator  14  having a port  14 A receiving, through transmission path  15 , output from port  12 B of the first circulator  12 . As seen by the relative position of ports  12 B and  12 A of the first circulator  12  along the circulator coupling direction  12 R, signals entering port  12 A couple to port  12 B. Likewise, as shown by their relative position along the second circulator coupling direction  14 R, signals entering port  14 A couple to port  14 B. Port  14 B of the second circulator  14  connects to the antenna feed port  14 . 
     Referring to prior art  FIG. 2 , it is readily seen that the circulators  12  and  14  form a directionally coupled transmit path for RF transmitter signals exiting the BST port  10  to reach the antenna feed port  16 , and form a directionally coupled reception path for reception signals to propagate from the antenna feed port  14 A to the input  18 A of the RF LNA  18 , and from the output  18 B of the RF LNA  18  to the BST port  10 . More specifically, with respect to the transmit path, a transmitter signal enters port  12 A of the first circulator  12 , leaves port  12 B, propagates along path  15 , enters port  14 A of the second circulator  14 , leaves port  14 B and, finally, enters the antenna feed port  16 . 
     With continuing reference to the prior art  FIG. 2 , as seen the selective coupling characteristic of the second circulator  14  substantially reduces transmitter signal entering port  14 A from coupling or otherwise leaking to port  14 C. This is significant, because such transmitter signal would otherwise enter the input  18 A of the RF LNA  18  and, hence, be amplified along with desired reception signal. 
     The above prior art methods apparatuses for TDD have various shortcomings. A significant shortcoming that is common to both these depicted switch-based and circulator-based TDD RF LNA isolators is that no failure-mode bypass of the RF LNA is provided. Methods for providing such bypass have been proposed and described, but each has various shortcomings and/or lacks features, benefits and advantages provided by exemplary embodiments and aspects according to the present invention. One illustrative example is the apparatus and method recited and illustrated at U.S. Pat. No. 6,812,786, issued 2004 to Jackson, et al. (hereinafter referenced as “Jackson &#39;786”). Jackson &#39;786 controls a selector switch 192 and controls two PIN diodes connecting to one hybrid 156, to selectively switch from the RF amplifiers 152 and 154 to the isolator 190. The selector switch 192 and its control are required, and as well as the isolator 190. 
     For these and other reasons the present embodiments provide advancements in the art, having various described features, advantages, and benefits as well as others that will be apparent to persons of ordinary skill in the art upon reading this disclosure. 
     SUMMARY 
     Various embodiments provide, among other features and benefits, a failure-mode bypass of the RF LNA without increase in noise figure (NF), with high switching speed, and without substantial increase in power dissipation, i.e., heat. 
     Various embodiments further provide, among other features and benefits, a minimal component count because of one or more aspects employing single components performing multiple substantial functions and, further, commonly structured components performing different functions, thereby providing performance benefits while, at the same time, reducing function-specific structure. 
     Applications of the various embodiments contemplate systems using RF amplifiers having a wide range of technologies and performance specifications and, therefore, for purposes of this description the term “RF amplifier” is defined as encompassing, but not being limited, to low noise amplifiers (LNA). 
     According to one exemplary embodiment, a system may have a TDD circuit for connecting a transceiver input/output (I/O) to an antenna, the circuit having a first hybrid coupler, having a first port connected or coupled to the antenna by, for example, a first path segment, and having a second hybrid coupler having a port connected, or coupled to the transceiver I/O by, for example, a second path segment. According to one or more aspects, a TDD circuit may have a first RF amplifier with an input connected by a first switchable impedance connector to a second port of the first hybrid coupler, and having an output connected by a second switchable impedance connector to a first port of the second hybrid coupler and, likewise, may have a second RF amplifier with an input connected by a third switchable impedance connector to the third port of the first hybrid coupler, and having an output connected by a fourth switchable impedance connector to a third port of the second hybrid coupler. A TDD circuit according to one or more aspects may further include a third hybrid coupler, having a first port connected to a fourth port of the second hybrid coupler, and having a second port connected to the fourth port of the first hybrid connector, a third port terminated by a first switchable impedance termination device, and a fourth port terminated by a second switchable impedance termination device. According to one or more aspects, each of the switchable impedance connectors, and each of the switchable impedance termination devices is configured to switch its impedance between a given matching impedance and a given mismatching impedance, in response to a given condition. 
     As will be understood, according to various aspects of one or more embodiments, a first hybrid coupler, a second hybrid coupler and third hybrid coupler may be arranged and configured to provide, in response to a concurrence of their switchable impedance connectors and a switchable impedance terminations having the mismatching impedance—resulting from an event such as, for example, a power supply failure—a bypass rearward directional path from the first port of the first hybrid coupler to the first port of the second hybrid device, the bypass rearward directional path bypassing the first and second RF amplifiers. 
     The above-summarized various illustrative examples of advances and advantages of the exemplary embodiments should not be understood as an exhaustive list or as a limit of the possible advantages that may be realized. On the contrary, various other advantages, as well as various alternative embodiments that are within the scope of the appended claims will be apparent to persons of ordinary skill in the art upon reading this disclosure. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  shows an exemplar according to a prior art switch-based TDD RF LNA isolation scheme; 
         FIG. 2  shows an exemplar according to a prior art circulator-based TDD RF LNA isolation scheme; 
         FIG. 3  depicts one illustrative example according to one TDD RF amplifier hybrid switch bypass of one or more embodiments; 
         FIG. 4  depicts one illustrative example implementation of one switching impedance termination using a PIN Diode of one example hybrid bypass switch aspect of one or more embodiments; 
         FIG. 5  illustrates an approximate equivalent circuit representing the  FIG. 3  example in its normal operating mode, during transmission of signal; 
         FIG. 6  illustrates an approximate equivalent circuit representing the  FIG. 3  example in its normal operating mode, during receiving of signal from the antenna; 
         FIG. 7  illustrates an approximate equivalent circuit representing the  FIG. 3  example in its bypass operating mode; and 
         FIG. 8  shows one example  800 , having embodiments as illustrated at  FIG. 3  and further including a receiving interval transmission path isolation aspect that disables the forward transmission path labeled  108  in  FIG. 3  during receiving intervals. 
     
    
    
     DETAILED DESCRIPTION 
     Various exemplary embodiments are described in reference to specific illustrative examples. The examples are included to further assist a person of ordinary skill in the art form a clear understanding of, and to practice the invention. The scope of embodiments, however, is not limited to the specific illustrative examples that are presented. Instead, as will be readily recognized by persons of ordinary skill in the relevant arts based on this description, many other configurations and arrangements may be implemented. 
     As will be appreciated by persons of ordinary skill in the art, for clarity of illustration figures may not be drawn to scale. For example, certain depictions may have distortion of shape and/or exaggeration of relative proportions, for purposes of a clear depiction of a whole. 
     To avoid obscuring novel features and aspects, and as will be readily understood by persons of ordinary skill in the art upon reading this description, various details of algorithms and hardware that are well known to such persons are omitted, except where such details pertain to particular operations of the features and aspects. 
     Example embodiments and aspects may be described separately, or as having certain differences. Separate description or description of differences, however, does not necessarily mean the respective embodiments or aspects are mutually exclusive. For example, a particular feature, function, or characteristic described in relation to one embodiment may be included in, or adapted for other embodiments. 
     In one example system according to one embodiment, a forward transmission path may be implemented by a first circulator located, for example, near the transmitter output, a first port of the first circulator receiving the forward signal and directionally coupling it to a second port, and two similarly coupled ports of a second circulator. The second circulator may be located, for example, near a feed port of the antenna. A forward transmission segment connects the second port of the first circulator to the first port of the second circulator, that port being directionally coupled to the second port of the second circulator. 
     In one example according to this and other embodiments, a receiving propagation path is provided from the antenna, in a rearward direction back to the RF receiving unit(s) of the transceiver. According to one aspect, the receiving propagation path includes one or more receiving RF amplifier coupler-bypass embodiments that, in overview, provide a novel switchable connection of one or more RF amplifier(s) in the receiving propagation path segment with, for example, two operating modes—a normal mode and a bypass mode. 
     According to one aspect, an RF amplifier coupler-bypass embodiment includes one or more RF multiport couplers, having switchable impedances connected to one or more of the coupler&#39;s ports. The ports are hereinafter referenced as “I/O” ports because, unless otherwise stated, the multiport couplers are symmetric. 
     In one or more aspects of the RF amplifier coupler-bypass embodiments, the switchable impedances have two impedance states, one of the states referenced herein as a given matching impedance, the other referenced as a given mismatching impedance. As will be understood by persons of ordinary skill in the art upon reading this disclosure, the terms “matching” and “mismatching” are in reference to the given impedance of the multiport RF couplers. Further to this aspect of the RF amplifier coupler-bypass embodiments, the RF multiport couplers are structured to provide a normal coupling mode in response to the impedance state of the connected switchable impedances being one of the matching and mismatching states, and to provide a bypass coupling in response to these switchable impedances being the other of matching and mismatching. 
     To facilitate a ready understanding of the various aspects and embodiments, example implementations are described wherein the RF multiport couplers are configured to provide the normal coupling mode in response to the matching impedance state, and to provide the bypass mode in response to the mismatching impedance state. However, this is only an example for purposes of illustration, and alternate implementations may be identified by persons of ordinary skill in the art. 
     According to one aspect of one or more RF amplifier coupler-bypass embodiments using RF multiport couplers with switchable impedances connected to one or more of the couplers&#39; I/O ports, the couplers are arranged to provide, in response to switching the impedances between the matching state and the mismatching state, a corresponding changing between a normal mode RF amplifier path—in which the RF amplifier(s) is(are) in line with the receiving propagation path—and a bypass mode RF amplifier path—in which the receiving propagation path bypasses the RF amplifier(s). 
     According to one or more aspects of at least one RF amplifier coupler-bypass embodiment using switchable impedances, the switchable impedances may receive a system power and, in response to the system power changing between a given acceptable level and a given failure level, may switch between the matching and the mismatching states. This system power may be common with a system power provided to the RF amplifier(s). Therefore, as will also be understood, among the various benefits and features of this aspect of the RF amplifier coupler-bypass embodiments is that, upon a failure of the system power that causes failure of the RF amplifiers, the switchable impedance switches, since the power failure causes these to switch their impedance state to the mismatching state which, in turn, causes the RF multiport couplers to provide the bypass path, the RE amplifier(s) are automatically bypassed in the event of a system power failure. 
     According to one aspect of one or more embodiments, the RF amplifier coupler-bypass circuit may include a transmit interval RF amplifier isolation aspect, operational during the normal mode (i.e., when power is available to the RF amplifiers) that, during transmit intervals, (meaning intervals in which the transmitter of the transceiver unit is transmitting signals to the antenna feed port) effectively isolates the RF receiver amplifiers from the antenna feed port. As will be understood from the detailed description, among other advantages and benefits, this provides a substantial reduction of the RF amplifiers, whose intended function is amplifying signals received from the antenna feed, from receiving transmitted signal energy during the transmit interval, (such as from imperfect circulators) and then amplifying such signals, and then inputting the amplified signal to the receiver of the transceiver unit. 
     For purposes of consistent referencing, this RF amplifier isolation aspect will be referenced hereinafter as “transmit interval RF amplifier isolation.” 
     According to one aspect of one transmit interval RF amplifier isolation embodiment, a transmitted signal power detection circuitry detects signal power generated by the transmitter of the transceiver unit, and generates a corresponding transmitter power detection signal. The transmitter power detection signal may be referenced as, for example, a TX/RX control signal. Further to the one aspect, isolation switches are controlled by the TX/RX control signal such that, in the absence of transmitter power, the RF amplifiers are switched into the return path from the antenna feed to the transceiver port (and thus to the RF receiver of the transceiver unit) and, in the presence of transmitter power, the RF amplifiers are switched out of the return path from the antenna feed to the transceiver port. 
     According to one example having one or aspect of RF amplifier coupler-bypass embodiments, the RF multiport couplers may comprise what will be termed “dual-mode” 90-degree hybrid couplers” each having an associated switchable impedance to switch from a normal coupling mode to a bypass coupling mode. According to one example, a dual-mode 90-degree hybrid coupler may comprise a conventional 90-degree hybrid coupler having a first I/O port, a second I/O port, a third I/O port and a fourth I/O port, with the first I/O port being the designated input port, and having a first switchable impedance connected to the third I/O port and a second switchable impedance connected to the fourth IO port. In one example, the dual-mode 90-degree hybrid coupler may be implemented by a 90-degree hybrid coupler structured and configured to provide a “normal mode” coupling from the first I/O port to the third and fourth I/O ports when the first and second switchable impedances have the matching impedance state and, in contrast, to provide a “bypass mode” coupling from the first I/O port to the second I/O port when the first and second switchable impedances have the mismatching impedance state. 
     According to one or more aspects of various RF amplifier coupler-bypass embodiments, an RF amplifier coupler-bypass circuit may be arranged in the reception propagation path, the circuit having a plurality of dual-mode hybrid phase shift couplers, arranged with one or more RF receiving amplifiers. In one example arrangement having one or embodiments, in response to a “normal mode” condition, a first of the dual-mode 90-degree hybrid couplers provides a propagation path to the input(s) of the RF receiving amplifiers, and a second of the dual-mode 90-degree hybrid couplers provides a propagation path from the output(s) of the RF receiving amplifier(s) to the feed or connection port of the RF transceiver. In example aspects having a circulator near the connection or feed of the transceiver, this propagation path includes respective ports of that circulator. 
     Further, according to various aspects of one or more exemplary embodiments, the above-described RE amplifier coupler-bypass circuit may be configured and arranged with a third of the dual-mode 90-degree hybrid couplers to provide, in response to a “bypass mode” condition, a bypass path around the RF receiving amplifiers, through these first, second and third dual-mode 90-degree hybrid couplers. As will be described in greater detail in later sections, in the event of, for example, a power failure that disables the RF receiving amplifier proximal to the antenna, this aspect automatically provides, despite the power failure, the RF reception signal to propagation to the RC receiver unit(s) within the system transceiver. 
     In one example having the transmit interval RF amplifier isolation aspect, an attenuation path may be arranged in parallel with each of the RF receiver amplifiers with, for example, an arrangement of 1:2 or 2:1 single-pole-double-throw (SPDT) switches arranged to switch according to the TX/RX control signal such that, in response to detected transmitted signal power, the RF amplifiers are switched out of the return path from the antenna feed to the transceiver port and the attenuators are connected in their place. 
     According to one or more of the isolation aspects, the 1:2 and 2:1 SPDT switches for switching between the RF receiver amplifiers and the attenuators may be constructed and arranged to additionally provide the switchable impedance function. In other words, according to the these aspects, in the 1:2 and 2:1 SPDT switches&#39; normal mode of operation, e.g., when the switches receive adequate power, in addition to their performing their respective receive and transmit mode switching, each switch also provides a matching impedance at the I/O port of dual-mode 90-degree hybrid couplers, such that the couplers operate in their normal mode. 
     As will be understood, among the various features and benefits of one transmit interval RF amplifier isolation aspect is that no external control signal is needed to control the isolation switches. On the contrary, the transmitted signal power detection circuitry simply detects when a transmitter of the transceiver unit is transmitting signal power to the antenna, generates the TX/RX or equivalent transmitter power detection signal and, in response, the isolation switches remove the RF amplifiers from return path from the antenna feed to the transceiver unit port. Among other features and benefits provided by this aspect is an asynchronous sharing of the antenna by one or more transmitters within the transceiver unit One among the example benefit according to one aspect is a hardware savings, as 90-degree hybrid couplers implementing the isolation aspect may be common with hybrid couplers bypassing the RF receiver amplifiers in the event of, for example, a power failure. 
     According to one or more of the various embodiments having the above-described first and second circulators, the forward transmission path may include a receiving interval transmission isolation According to various aspects, receiving interval transmission isolation enable the forward transmission path during a transmission interval (i.e., when the transmitter power detection signal or TX/RX control signal indicates signals are being transmitted), and disable the forward transmission path outside of the transmission interval. In the TX mode the isolation circuit provides a low loss path of less than 0.5 dB. In the RX mode it attenuates the path by 12 to 18 dB Further, according to various exemplary embodiments the receiving interval transmission isolation includes a normal/bypass aspect that disables the isolation in the event of a failure condition such as, for example, the above-described failure to receive a normal power at the RF receive amplifier (and at the switchable impedances within the RF amplifier coupler-bypass circuit). According to one or more aspects, the isolation and the disabling of the isolation may be implemented by, for example, a dual-mode 90-degree hybrid coupler having switchable impedances connected to two of its designated I/O ports. 
       FIG. 3  illustrates one example TDD Hybrid Bypass system  100  having various aspects of one or more of the exemplary embodiments. It will be understood that the system  100  is depicted as functional blocks, each block representing circuit transfer functions, connected by lines representing signal paths. Unless otherwise stated, or made otherwise clear from the particular context, the technology and performance specifications of signal paths connecting functional block are not particular to the invention, and may implemented using conventional design methods and conventional technology, based on selection criteria ordinarily applied by persons of ordinary skill in the arts pertaining to the invention. It will also be understood that signal paths represented by the interconnecting lines on  FIG. 3  are not necessarily implemented as simple transmission paths and may, for example, include devices such as limiters, amplifiers and filters, in accordance with convention design and performance considerations. 
     Referring now to  FIG. 3 , the example system  100  includes a BTS port  102  that may, for example, feed an output of a transmitter (not shown) and an input of a receiver (not shown). The BTS port  102  connects via a transmission path  104  to an “A” port  106 A of a first circulator  106 . The example first circulator  106  has a second signal port  106 B and a third port  106 C. It will be understood that the port labels of “A,” “B,” and “C” are arbitrary. The principles of operation, available structures, and circuit considerations for using circulators are known to persons of ordinary skill in the art and, therefore, except for purposes of describing their interface within the example systems described herein, further detailed description is omitted. Further, persons of ordinary skill in the art, upon reading the present disclosure in its entirety, can readily specify performance criteria the circulator must meet, by applying conventional RF design considerations to the particular system specifications and its the particular application. 
     With continuing reference to  FIG. 3 , the directed arrow  106 R represents the coupling characteristic of the first circulator  106 . As shown, transmission signals that enter I/O port  106  will, through a directional coupling mechanism forming, from port A and port B, a forward transmission path as known to persons of ordinary skill in the art, exit port  106 B. A forward transmission path  108  connects port  106 B of the first circulator  106  to port  110 A of a second circulator  110 . The  FIG. 3  example  100  shows the example forward transmission path  108  as a simple transmission path, with no switching structure. The depicted path  108  is, as will described in greater detail in later sections and in reference to later figures, other aspects of these and other exemplary embodiments may include one or more directional couplers (not shown in  FIG. 3 ) and, further, may include a receiving interval transmission isolation aspect (not shown in  FIG. 3 ) that, for example, isolates the LNA signals from being fed back upon itself through the TX path during the receiving interval. 
     Referring to  FIG. 3 , I/O port  110 B of the second circulator  110  may connect, in the depicted example  100 , through a bandpass filter  112  to an antenna feed port  114 . Specifications of the bandpass filter  112  and the technologies for its implementation, as well as the factors and considerations for including or omitting such a filter may be according to conventional transceiver design, and are readily identifiable to persons of ordinary skill in the art based on this description. Further detailed description of the bandpass filter  112  is therefore unnecessary for understanding and practicing the embodiments and, thus, is omitted. 
     With continuing reference to  FIG. 3 , port  110 C of the second circulator  110  connects through a path (not separately labeled) to input  116 A of multiport coupler  116 . The example arrangement of the multiport coupler  116 , together with the multi-port couplers  128  and  136 , and their associated example circuitry provide, among other functions and features, a “normal” mode, and a “bypass” mode of operation. As will be understood, in the normal mode the RF amplifiers  122  and  132  are operational and, during the receiving time interval (i.e., the time interval when no transmitted signals are output from the BTS port  102  and the antenna is allocated to receiving signals) the RF amplifiers  122  and  132  are switched, by actions of the multiport couplers  116 ,  128  and  136  that are described in greater detail in later sections, to be within the path from the antenna feed port  114  back to the BTS port  102 . In the bypass mode, on the other hand, such as may be occur, for example, in response to a power failure disabling the RF amplifiers  122 ,  132 , later-described actions of the multi-port couplers  116 ,  128  and  136  and their supporting circuitry, provide a bypass path from the antenna feed port  114  back to the BTS port  102  that does not include the RF amplifiers  122 ,  132 . 
     Referring still to  FIG. 3 , the example system  100  also includes one example implementation of a transmit interval RF amplifier isolation aspect as described previously. As described previously, and as will be described in greater detail in later sections, the  FIG. 3  transmit interval RF amplifier isolation aspect includes a transmitted power detector  107  that detects transmitted signal power on the forward path  108  and based, for example, on a comparison to a given threshold, generates a corresponding transmit/receive mode control signal TX/RX. The TX/RX control signal may be received by switching circuitry, described in greater detail in later sections, that during the detected receiving interval (i.e., no communication signal output from the BTS port  102 ) maintains the RF amplifiers  122 ,  132  in the reception path from the antenna feed  114  to the BTS port  102  and, during the transmit interval, isolates the RF amplifiers  122 ,  132  from any path from the antenna feed  114  to the BTS port  102 . As previously described, this isolation prevents the RF amplifiers  122 ,  132  from receiving coupled and/or radiated energy of the transmitted signal, amplifying it, and returning the amplified signal back to the BTS port  102 . 
     With continuing reference to  FIG. 3 , in the depicted example  100 , the transmit interval RF amplifier isolation aspect is provided by the switchable impedances  118 ,  120 ,  124  and  134  being configured to provide, in addition to the switchable impedance, the functions of a 1:2 or 2:1 SPDT switch. The example switchable impedances  118 ,  120 ,  124  and  134 , because of including the 1:2 or 2:1 SPDT function, are therefore referenced hereinafter as “switchable impedance SPDT switches.” It will be understood, though, that the “SPDT switch” feature is optional, as embodiments may omit the transmit interval RF amplifier isolation feature. As will be further described in later sections, in the example  100  a transmit-interval transmission isolation control such as, for example, the depicted power detector  107  may detect transmitter power in, for example, the transmission path  108  and, in response, may generate a control signal (not shown in  FIG. 3 ) that is received by each of the switchable impedance SPDT switches  118 ,  120 ,  124  and  134 . When the power detector  107  detects transmitter power in the transmission path  108 , SPDT switches  118  and  120  switch to connect their respective “C” inputs to their respective B” outputs, and SPDT switches  124  and  134  switch to connect their respective “B” inputs to their respective “C” outputs. As a result, transmitter power leaking or coupling from transmission line  108 , or from reflections from the antenna (not shown in  FIG. 3 ) is substantially prevented from entering inputs (not separately numbered) of the RF amplifiers  122  and  132 . Instead, such transmitter energy may dissipate in the attenuators  126 ,  130 . To facilitate a ready understating of the embodiments, description of example aspects and operations of the multi-port couplers  116 ,  128  and  136  will first assume that the SPDT switches  118 ,  120 ,  124  and  134  are in their transmit isolation mode, i.e., that the system  100  is operating in a reception interval. 
     Referring to  FIG. 3 , in the particular arrangement depicted in the example  100 , the multiport couplers  116 ,  128  and  136  will be assumed to be 90-degree hybrid couplers. This assumption does not, however, limit the practicing of various embodiments illustrated by the  FIG. 3  example  100  to using a 90-degree hybrid coupler. On the contrary, other implementations of the embodiments and aspects contained in the example  100  may be apparent to persons of ordinary skill in the art, by a corresponding arrangement of other couplers. For purposes of readily describing example operations on the example  100 , though, the example implementation of the multiport coupler  116  is a 90-degree hybrid coupler and, therefore, unit  116  will be referenced as a “90-degree hybrid coupler”  116 . 
     Referring again to  FIG. 3 , example aspects and operations of each of the multiport couplers  116 ,  128  and  136  having two operative modes of coupling, one being the above-described “normal mode,” and the other being the above-described “bypass” mode, and how these combine to provide the overall normal and bypass mode will be described. In one or more example embodiments, each of the 90-degree hybrid couplers  116 ,  128  and  136  has switchable impedances connected to at least two of its I/O ports. With respect to the 90-degree hybrid couplers  116 , the switchable impedances are the SPDT switches  118  and  120  connected to its ports  116 C and  116 D. With respect to the 90-degree hybrid coupler  128 , the switchable impedances are the SPDT switches  130  and  134  connected to its ports  128 A and  128 B. Lastly, with respect to the 90-degree hybrid coupler  138 , the switchable impedances are the switchable impedances  138  and  140  connected, respectively, to ports  138 C and  138 D. Example implementations of the switchable impedances  138  and  140  are described in greater detail in later sections. 
     With continuing reference to  FIG. 3 , the SPDT switches  118  and  120  connected to ports  116 C and  116 D, switch the hybrid coupler  116  between its normal and bypass modes of operation by switching between two input impedance states; one matching (within a given tolerance) the given characteristic impedance of the 90-degree hybrid coupler  116 , and the other state being a given mismatch relative to that given characteristic impedance. The term “input impedance,” with respect to the example switches  118  and  120 , means the impedance looking into their respective “c” input ports. The normal mode of coupling, resulting from the input impedances at the c inputs of  118  and  120  having their matching state, will be described first. 
     As a preliminary matter, as known to persons or ordinary skill in the art, any I/O port of a four-port 90-degree hybrid may be chosen as an “input,” and, if matching impedances are connected to the remaining three ports, the hybrid exhibits the following coupling characteristic: The input signal is split within the coupler, in a power splitter manner, and two 3 dB attenuated versions of the signal are output from a respective two of the other four I/O ports. One of these two 3 dB attenuated signals has a given reference latency (which may be zero degrees), while the other of the two 3 dB attenuations of the signal has a 90-degree phase delay relative to the signal output from the first of the two “output” ports. Propagations of the input signal, however, cancel at the fourth I/O port. 
     In view of this coupling behavior of a 90-degree hybrid, and referring to example  100  of  FIG. 3 , when the switches  118  and  120  having the matching impedance at their respective inputs  118 C and  120 C, the hybrid coupler  116  acts as a power splitter, outputting, in response to a signal input at port  116 A, a 3 dB attenuation of the input signal from port  116 D and a 3 dB attenuation, with a 90-degree delay, from port  116 C. To best provide a consistent terminology in this description, the terms “on-phase” and “quadrature” will be used to identify, in reference to the power splitter mode of the 90-degree hybrids, the 3 dB attenuated signal having a “zero” phase delay and the 3 dB attenuated signal having a 90-degree phase delay. The term “zero” is a relative term, as structural aspects of a 90-degree hybrid may introduce certain delay common to both the in-phase and quadrature signal paths. 
     Continuing with the above-describer illustrative “normal” operation of the 90-degree hybrids  116 , i.e., with all of the above-described switchable impedance at their matching state, a reception signal from the antenna feed  114  passes through the BPF  112 , into port  110 B and out from port  110 C of the second circulator  110 . For purposes of reference, the reception signal will be arbitrarily labeled as RC(t). This RC(t) enters port  116 A of the hybrid  116  and, as a result, a 3 dB attenuated in-phase RC(t) exits port  116 D and, assuming switch  120  is in the receiving mode, the 3 dB attenuated in-phase RC(t) is input to the RF amplifier  132 . Likewise, a 3 dB attenuated quadrature RC(t) is output from port  116 C and input to the RF amplifier  122 . Assuming the RF amplifiers  122  and  132  have a 30 dB gain, a 27 dB amplified in-phase RC(t) is output from amplifier  122  and input to port  128 B of the second 90-degree hybrid  128 , and a 27 dB amplified quadrature RC(t) is output from amplifier  132  and input to port  128 A of the second 90-degree hybrid  128 . The propagation through the second 90-degree hybrid  128  will now be described. With respect to the 27 dB in-phase signal input to port  128 A, a 3 dB attenuated in-phase RC(t) will arrive at port  128 C, and a 3 dB attenuated quadrature RC(t) will arrive at port  128 D. With respect to the 27 dB quadrature RC(t) that is input to port  128 B, a 3 dB attenuated in-phase RC(t) will arrive at port  128 D, and a 3 dB attenuated quadrature RC(t) will arrive at port  128 C. The total delay relative to RC(t) entering port  116 A of the first 90-degree hybrid  116  for the quadrature of RC(t) propagating from port  128 A to port  128 D is therefore minus 180 degrees, while the total delay of the in-phase signal propagating from port  128 B to port  128 D is zero degrees. Therefore, omitting phase differences these signals cancel. On the other hand, referring to port  128 C of the second hybrid coupler  128 , the RC(t) it receives from port  128 A has the same phase delay as the RC(t) it receives from port  128 B. The signals therefore add, producing a 27 dB gain RC(t) that, in the  FIG. 3  example  100 , is input to port  106 C of the first circulator  106  and then, according to the depicted characteristic of the circulator  116 , out from port  106  A to the BST  102 . 
       FIG. 4  shows one example implementation  400  of the switchable grounding impedances  138 ,  140  connected to ports  136 C and  136 D of the 90-degree hybrid  136 . Referring to  FIG. 4 , the example  400  comprises a PIN diode  402  fed by a bias generator (not shown) outputting a two voltage state signal (not shown in  FIG. 4 ) such as, for example, +5 and −70 volts. 
       FIG. 5  illustrates an approximate equivalent circuit representing the  FIG. 3  example  100  in its normal operating mode, during transmission of signal into port  106 A of circulator  106 . Referring to  FIGS. 3 and 5  together, the switchable impedance SPDT switches  118  and  120  are in their matched impedance state, and are each switched to connect their respective “C” inputs to their respective “B” outputs. Further, the switchable impedances  138 ,  140  connected to ports  136 C and  136 D of the 90-degree hybrid  136  provide proper matched impedance and, therefore ports  136 A and  136 B provide match impedance for ports  116 A of the 90-degree hybrid  116  and port  128 C of the 90-degree hybrid  128 . All of the ports of the 90-degree hybrid  116  and  128  are therefore terminated by matched impedances. Accordingly, each of the 90-degree hybrids  116  and  128  operate in their normal mode, each functioning as a power splitter. As shown, any transmission signal entering port  116 A of the 90-degree hybrid  116  is split into an attenuated in-phase and 90-degree delayed component output, respectively, from ports  116 C and  116 D of the 90-degree hybrid  116  and dissipated by the attenuators  126 ,  134 . The attenuators  126  and  134  are readily specified and implemented by a person or ordinary skill in the art, based on this disclosure, such that any leakage or coupling energy of the transmitter signal is sufficiently attenuated by the devices  126 ,  134 . 
       FIG. 6  illustrates an approximate equivalent circuit representing the  FIG. 3  example  100  in its normal operating mode, during a receiving of signal from the antenna (not shown in  FIG. 3 ). Comparing  FIG. 6  to  FIG. 5 , it is seen that switchable impedance SPDT switches  118  and  120  connect their respective “C” inputs to their respective “A” outputs which, in turn, are connected to the inputs  122 A and  132 A or the RF amplifiers  122 ,  132 . In a corresponding manner, SPDT switches  124  and  134  connect their respective “A” inputs, each receiving a corresponding one of the outputs of the RF amplifiers  122 ,  132 , to the switches respective “C” output. Reception signal entering I/O port  116 A of the 90-degree hybrid  116  is split into the above-described two signals, one being an in-phase signal exiting I/O port  116 D, the other being a 90-degree delayed signal exiting port  116 C. These signals are input, respectively, to the RF amplifiers  122  and  132 , and each is amplified by, for example, 30 dB. Proceeding with the description in the direction of the example reception signal propagation, an amplified in-phase reception signal from the output  132 B of the RF amplifier  132  passes through the switchable impedance SPDT switch  134  and into I/O port  128 B of the 90-degree hybrid coupler  128 . Similarly, an amplified 90-degree delayed reception signal output from the output  122 B of the RF amplifier  122  passes through the switchable impedance SPDT switch  124  and into I/O port  128 A of the 90-degree hybrid coupler  128 . 
     With continuing reference to  FIG. 3 , as readily seen and understood by a person of ordinary skill in the art in view of this disclosure, the 90-degree hybrid coupler  128  splits the amplifier 90-degree delayed amplified reception signal entering its port  128 A into two signals. Likewise, the 90-degree hybrid coupler  128  splits the amplifier 90-degree delayed amplified reception signal entering its port  128 A into two signals. The signal arriving at port  128 C from port  128 A is the quadrature signal from port  128 A and, since the signal entering port  128 A is in-phase relative to the input at  116 A, the total delay is minus 90-degrees. The signal arriving at port  128 C from port  128 B is the in-phase signal from port  128 B but, since the signal entering port  128 B is the quadrature signal relative to the input at  116 A, the total delay is the same as the total delay of the signal arriving at port  128 C from port  128 A, namely minus 90-degrees. Therefore, the signal arriving at port  128 C from port  128 A adds to the signal arriving at port  128 C from port  128 B. In contrast, the signal arriving at port  128 D from port  128 A cancels, assuming ideal operation by the 90-degree hybrids  116  and  128 , the signal arriving at port  128 C from port  128 B. 
     As can be seen from  FIG. 3 , the signal arriving at port  128 D from port  128 A is in-phase relative to port  128 A, and the signal entering port  128 A is in-phase relative to the signal entering port  116 A. Thus, with respect to the signal entering port  128 D from port  128 A, the overall phase delay is zero. On the other hand, the arriving at port  128 D from port  128 B is the quadrature signal from port  128 B and, since the signal entering port  128 B is the quadrature signal relative to the input at  116 A, the total delay is minus 180 degrees. Therefore, the signal arriving at port  128 D from port  128 A cancels the signal arriving at port  128 D from port  128 B. 
     As persons of ordinary skill in the art will understand from this disclosure, factors such as different path delays may result in phase difference between the two signals arriving at port  128 C, as well as two signals arriving at port  128 D. A distortion and attenuation may result. However, persons of ordinary skill in the art can, based on given system requirements, readily identify bounds for such factors, and applying conventional know-how to this disclosure, can readily select circuit topologies and components accordingly. 
       FIG. 7  illustrates an approximate equivalent circuit representing the  FIG. 3  example  100  in its bypass operating mode. 
     Referring to  FIGS. 6 and 7 , it is assumed that the switchable impedance switches  118 ,  120 ,  124  and  134  are all in their mismatching impedance state such as, for example, would result from a power supply failure. Likewise, it is assumed the switchable impedance terminations  138  and  140  are in their mismatching state. The result is that I/O port  116 B forms a direct path to I/O port  116 , I/O port  128 A forms a direct path to I/O port  128 B. Similarly, I/O port  136 A forms a direct path to I/O port  136 B. 
       FIG. 8  shows one example  800 , having embodiments as illustrated at  FIG. 3  and further including a receiving-interval transmission path isolation aspect that disables the forward transmission path labeled  108  in  FIG. 3  during receiving intervals. Referring to  FIG. 8 , in the example  800  the illustrative implementation of the receiving-interval transmission path isolation aspect is a 90-degree hybrid  802 , fed at its I/O port  802 A by transmission signal from I/O port  106 B of the first circulator  106 , having switchable impedances  804  and  806  terminating hybrid I/O ports  802 C and  802 D, and I/O port  802 B of the 90-degree hybrid  80  feeding I/O port  110 A of the second circulator  110  through transmission path  808 . The 90-degree hybrid  802  may, for example, be structurally identical to a 90-degree hybrid implementing the hybrid coupler  136  and, likewise, the switchable impedances  804 ,  806  may be structurally identical to the switchable impedances  138 ,  140  terminating I/O ports  136 C,  136 D, respectively, of the coupler  136 . As described above, and as shown by the PIN diode example  400  shown in  FIG. 4 , in the event of a given condition such as, for example, a power failure, the switchable impedances  138 ,  140  assume their mismatching impedance state. Therefore, in an example implementation of the  FIG. 8  example  800  that uses the same or equivalent structure for the switchable impedances  804 ,  806  as for the impedances  138 ,  140 , the same given event, namely a power failure, causes the impedances to  804 ,  806  to assume the mismatching impedance state. The result is that, with respect to transmission signal entering port  802 A, the ports  802 C and  802 D will reflect the in-phase and quadrature signals arriving at these ports (i.e., the respective signals that these ports receive as a result of the signal entering port  802 A) back into the ports  802 C,  802 D, and these reflected signals will sum at port  802 B. The summed signal then propagates through path  808  to the I/O port  110 A of the circulator  110 . In other words, in the event of, for example, a power failure the 90-degree hybrid  802  acts as a straight through coupler between its ports  802 A and  802 C. 
     With continuing reference to  FIG. 8 , illustrative example operations of the receiving-interval transmission path isolation aspect will be described. Unless otherwise stated, described example operations assume a normal state, e.g., an acceptable power supply level. Referring to  FIG. 8 , a transmission interval TI may be defined as the interval during which transmission signal power is detected as output from port  106 B of the circulator  106 . In the  FIG. 8  example, the detection is performed by a power detector  808 . The power detector  808  may be the previously described power detector  107 . Resulting from the detection the power detector  808  outputs the previously described transmit/receive signal TX/RX having an appropriate value, arbitrarily labeled as TR that, in accordance with the  FIG. 8  arrangement, causes the switchable impedances  804 ,  806  to assume their mismatching state. Assuming the switchable impedance  804 ,  806  are PIN diodes as illustrated at  FIG. 4 , the TR value is determined by the voltage needed to reverse bias the PINs which may, for example, be minus 75 volts. As a result of the mismatching impedance of 90-degree hybrid  802  acts as a straight-through coupler between its ports  802 A and  802 C. Therefore, during the transmission interval TI, transmission signal enters I/O port  106 A of the first circulator, exits the circulator port  106 B, propagates into I/O port  802 A and out of I/O port  802 B of the 90-degree hybrid  802 , then through path  808 , into I/O port  110 A of the second circulator  110  and, finally, from I/O port  110 B of the second circulator  110  through the BPF  112  (if included) and into the antenna feed port  114 . As readily seen, during intervals in which the power detector  808  does not detect transmission signal power, i.e., during receive intervals RI, the detector  808  generates a TX/RX signal having an appropriate value RR that causes the switchable impedances  804 ,  806  to assume their matching state. As described above, one illustrative matching state impedance is 50 ohms, at the signal frequency at which the system operates. Continuing with the assumption that the switchable impedances are PIN diodes as shown at  FIG. 4 , one illustrative example value RR is positive five volts. Upon receiving a five volt TX/RX, the switchable impedances  804 ,  806  assume their matching state and, as result any transmission signal energy entering the I/O port  802 A exits I/O ports  802 C,  802 D as an in-phase and quadrature signal, respectively. Since the switchable impedances  804 ,  806  are in their matching impedance such signals are grounded. 
     As shown in the example  800 , a directional coupler such as item  810  may be inserted in the path between the I/O port  106 B of the circulator  106  and the I/O port  802 A of the 90-degree hybrid. The directional coupler could also be inserted before the first circulator  106 . The purpose and function of the directional coupler  810  is to prevent, or reduce to an acceptable level, energy reflected back from the I/O port  802 A from travelling back to port  106 B of the circulator  106 . It will be understood that the directional coupler  810  may be used as well in the forward transmission path  108  of the  FIG. 3  example  100 . 
     The above-described embodiments and aspects provide various advantages, benefits and features that constitute significant advances over the prior art, in terms of, for example, scope of functions, performance, reliability and testability. 
     One among the various examples is provided by the hybrid coupler  136 , in its depicted arrangement of connecting to the 90-degree hybrid coupler  128  and to the 90-degree hybrid coupler  116 , the 90-degree hybrid coupler  136  itself terminated with controllable impedances  138  and  140 . This novel arrangement and structure provides for the 90-degree hybrid coupler  136  to server two functions; one function being the proper termination impedance for the 90-degree hybrid couplers  128  and  116  when proper supply power is available (and hence the RF amplifiers  122 ,  132  function), and the other function being part of the bypass path occurring n the event of, for example power failure (by way of the path from  136 A to  136 B resulting from impedances  138  and  140  switching to their mismatched state). 
     Another among the various examples is provided by the transmit-interval RF amplifier isolation aspect, described in reference to  FIGS. 3-8 , that, by using the power detector  107  (or  808 ) generates the TX/RX control signal to switch the RF amplifiers  122 ,  132  in and out of the return path, without requiring external control signals. 
     Another among the various example advantages, benefits and features is provided by the receiving-interval transmitter isolation aspect, e.g., the isolation hybrid coupler  802  described in reference to  FIG. 8  that, in the transmit mode (i.e., when the power detector  808  generates the TX/RX signal to set the impedances  804 ,  808  to their mismatch state) provides a low-loss forward path but, during the receiving interval (i.e., when the power detector  808  generates the TX/RX signal to set the impedances  804 ,  808  to their matching state), provides high isolation between the antenna feed port  114  and the BTS port  102 . One particular contemplated advantage of this is attenuation of out-of-band reflections from the LNA in the RX mode from the band pass filter  112 . 
     Another among the various example advantages, benefits and features is provided by the 1:2 and 2:1 SPDT features of the FET switches  118 ,  120 ,  124  and  134  that provide selecting a path through the RF amplifiers  122 ,  132  in the receiving mode and a path through the attenuators  126 , 130  in the transmit mode. As will be understood by a person of ordinary skill in the pertinent art upon reading this disclosure, embodiments combining the receiving-interval transmitter isolation aspect, e.g., the isolation hybrid coupler  802  and the 1:2 and 2:1 SPDT features of the FET switches  118 ,  120 ,  124  and  134  providing a selectable a path through the RF amplifiers  122 ,  132  in the receiving mode, and through the attenuators  126 , 130  in the transmit mode, permit introduction of a calibration signal into the BTS port  102  in the receive mode only and read back through the RF amplifiers  122 ,  132 . Prior art TDD systems using switches cannot be calibrated in this manner. 
     As will also be understood by persons skilled in the arts, upon reading this disclosure, this combination even further allows measurement of return loss in the transmit mode. 
     Although the various exemplary embodiments have been described in detail with particular reference to certain exemplary aspects thereof, it should be understood that the invention is capable of other embodiments and its details are capable of modifications in various obvious respects. As is readily apparent to those skilled in the art, variations and modifications can be affected while remaining within the spirit and scope of the invention. Accordingly, the foregoing disclosure, description, and figures are for illustrative purposes only and do not in any way limit the invention, which is defined only by the claims.