Abstract:
Disclosed is a mixer able to simultaneously suppress self-mixing and low-order harmonic response in a charge sampling circuit. Specifically disclosed is a multiphase mixer provided with a transconductance amplifier ( 101 ) for converting a voltage signal into a current signal, an N number (where N is a natural number that is 2 or more) of first integrators ( 401, 402 ) which are connected in parallel to the subsequent stage of the transconductance amplifier ( 101 ), and a 2N number of mixers ( 102, 103, 104, 105 ) connected in parallel in pairs to the respective N number of first integrators ( 401, 402 ), wherein two mixers connected to the same first integrator of any of the N number of first integrators ( 401, 402 ) are controlled by driving signals comprised of pulse trains with the same frequency and phases differing by 180°.

Description:
TECHNICAL FIELD 
     The present invention relates to a self-mixing suppression technology for a mixer used in a high frequency processing section in a radio communication system, and a mixer harmonic response suppression technology. 
     BACKGROUND ART 
     A TV tuner needs to cover a wide reception band allocated to TV broadcast signals. For example, in Japan, a TV tuner needs to support VHF (Very High Frequency) channels (100 MHz band, 200 MHz band) and UHF (Ultra High Frequency) channels (470 MHz to 770 MHz). Also, a software radio needs to support a plurality of radio systems that use different radio bands. 
     In general, when a radio frequency signal of a frequency band that is an odd multiple of a local oscillation signal for driving a mixer is input to a mixer configuring a radio reception section, a disturbing signal frequency-converted to a frequency in the vicinity of received signal output having a desired frequency is output due to a nonlinear characteristic of the mixer. Also, in general, it is possible to suppress a harmonic response that is an even multiple of a local oscillation signal by giving a mixer a differential configuration, and using differential combining in subsequent-stage circuitry. However, if a radio frequency signal of a frequency band that is an even multiple of a local oscillation signal is input due to a mismatch between differential circuits, a disturbing signal frequency-converted to a frequency in the vicinity of received signal output having a desired frequency is output. Below, this disturbing signal is referred to as harmonic response. 
     Here, if a reception band that should be supported by a TV tuner or software radio is wide, and the ratio between signal amplitude when a signal of a desired frequency is received and signal amplitude of other than a desired frequency component output due to harmonic response reaches a predetermined value, reception sensitivity degrades. Thus, technology is known that suppresses harmonic response by approximating an output waveform to a sine wave (see Patent Literature 1 through Patent Literature 6, and Non-Patent Literature 1 and Non-Patent Literature 2). 
     Also, in recent years, with the object of simplifying the circuitry of a radio reception section, a direct conversion reception method whereby the output frequency of a high frequency processing mixer is set in the vicinity of zero hertz, or a Low-IF (Intermediate Frequency) reception method, have become mainstream. With these configurations, a local oscillation signal leaks to an RF input terminal of a mixer, this leakage component is reflected by a preceding-stage circuit and is input to the mixer again, and a DC component is output from the mixer. It is known that reception quality degrades due to fluctuation of the amplitude of this DC component. Below, this phenomenon is referred to as self-mixing. 
     In general, a reflection coefficient between a Low Noise Amplifier (LNA) connected ahead of a mixer and the mixer is changed by changing the LNA gain setting. Also, a reflection coefficient between an antenna and subsequent-stage circuitry changes according to a change in conditions around the antenna. The amplitude of a leakage component of a local oscillation signal fluctuates due to changes in these reflection coefficients. As a result, the amplitude of a DC (Direct Current) component fluctuates, causing degradation of reception sensitivity (self-mixing). A technology for suppressing this self-mixing is known whereby the ratio of an on-period to one cycle of a mixer driving signal (hereinafter referred to as the duty ratio) is made 25% (see Non-Patent Literature 3). 
     Also, in recent years there has been a technology called charge sampling, and charge sampling circuit  10  having a configuration such as shown in  FIG. 1  is known (see Non-Patent Literature 4, for example). Switch  2  is subjected to on/off control by means of a control signal comprising a rectangular pulse input from control terminal  11 , and a current output from current source  1  charges capacitative element  3  only while switch  2  is on. 
     A filter characteristic due to a current integration effect is obtained according to this charging period. Here, it is known that, among attenuation pole frequencies, a frequency closest to zero hertz changes according to the control signal duty ratio. Specifically, when the duty ratio is made N %, an attenuation pole is generated at a frequency position 100/N times the control signal (local oscillation signal) frequency. 
     CITATION LIST 
     Patent Literature 
     PTL 1 
     
         
         U.S. Pat. No. 3,962,551 specification
 
PTL 2
 
         U.S. Pat. No. 5,220,607 specification
 
PTL 3
 
         Japanese Patent Application Laid-Open No. SHO55-095178
 
PTL 4
 
         Published Japanese Translation No. 2005-536099 of the PCT International Publication
 
PTL 5
 
         Published Japanese Translation No. 2007-535830 of the PCT International Publication
 
PTL 6
 
         International Publication No. 2008/032782
 
PTL 7
 
         Japanese Patent Application Laid-Open No. 2004-289793 
       
    
     Non-Patent Literature 
     NPL 1 
     
         
         R. Bagheri, et al, “An 800 MHz to 5 GHz Software-Defined Radio Receiver in 90 nm CMOS”, Dig. Tech. Papers of the 2006 IEEE International Solid-State Circuits Conference (ISSCC), February, 2006, pp. 480-481.
 
NPL 2
 
         Weldon, J. A. et al, “A 1.75 GHz Highly-Integrated Narrow-Band CMOS Transmitter with Harmonic-Rejection Mixers”, Section 10.4 of Dig. Tech. Papers of the 2001 IEEE ISSCC, Feb. 5-7, 2001, pp. 160-162.
 
NPL 3
 
         Petrov, A. R., “System approach for low 1/f noise, high IP2 dynamic range CMOS mixer design,” University/Government/Industry Microelectronics Symp., 2003. Proc. of the 15th Biennial, Jun. 30-Jul. 2, 2003, pp. 74-77.
 
NPL 4
 
         Gang XU, et al, “Comparison of Charge Sampling and Voltage Sampling”, Proc. of the 43rd IEEE Midwest Symp. on Circuits and Systems, Aug. 8-11, 2000, pp. 440-443. 
       
    
     SUMMARY OF INVENTION 
     Technical Problem 
     With the charge sampling circuit configuration shown in Non-Patent Literature 4, it is considered to be possible to suppress self-mixing if the duty ratio is made 25%, as in Non-Patent Literature 3. Here, there is a problem in that, although an attenuation pole obtained through a current integration effect of a charge sampling configuration can be used for suppression of harmonic response, if the duty ratio is set to a value of less than 50%, such as 25%, an attenuation pole of twice the local oscillation signal generated when the duty ratio is 50% is shifted toward a higher-order frequency with respect to the local oscillation signal, with the result that low-order harmonic response can no longer be suppressed. Thus, with a charge sampling circuit configuration, suppression of self-mixing and suppression of low-order harmonic response close to a desired frequency band cannot be achieved simultaneously. 
     It is an object of the present invention to provide a mixer capable of simultaneously achieving suppression of self-mixing and suppression of low-order harmonic response with a charge sampling circuit configuration. 
     Solution to Problem 
     A multiphase mixer of the present invention, firstly, has a configuration comprising: a transconductance amplifier that converts a voltage signal to a current signal; N (where N is a natural number of two or more) first integration elements connected in parallel to a subsequent stage of the transconductance amplifier; and 2N mixers connected in parallel in pairs to the respective N first integration elements; wherein two mixers connected to the same integration element of any of the N first integration elements are controlled by driving signals comprising pulse trains having the same frequency and phases differing by 180°. 
     By means of this configuration, suppression of mixer harmonic response can be achieved. 
     A multiphase mixer of the present invention, secondly, is a multiphase mixer having the first configuration, having a configuration wherein each of the driving signals controlling the 2N mixers has an on-period ratio of 100/2N (%) with respect to one cycle. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, thirdly, is a multiphase mixer having the first or second configuration, having a configuration wherein the 2N mixers are controlled by driving signals comprising pulse trains having the same frequency and phases differing by 180/N(°), and of the 2N mixers, N mixers controlled by driving signals comprising pulse trains whose on-periods are adjacent are connected to the same second integration element. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, fourthly, is a multiphase mixer having the first or second configuration, having a configuration wherein the 2N mixers are controlled by driving signals comprising pulse trains having the same frequency and phases differing by 180/N(°), and of the 2N mixers, two mixers controlled by driving signals comprising pulse trains whose on-period phase difference is 180° are connected to the same second integration element. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, fifthly, is a multiphase mixer having any one of the first through fourth configurations, having a configuration wherein the 2N mixers are controlled by driving signals comprising pulse trains that are not in an on state simultaneously. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, sixthly, is a multiphase mixer having any one of the first through fifth configurations, having a configuration wherein the N is 2. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, seventhly, is a multiphase mixer having any one of the first through fifth configurations, having a configuration wherein the N is 3. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, eighthly, has a configuration comprising: a transconductance amplifier that converts a voltage signal to a current signal; a first integration element connected in parallel to a subsequent stage of the transconductance amplifier; and M (where M is a natural number of two or more) mixers connected in parallel with respect to the first integration element, and controlled by driving signals comprising pulse trains having the same frequency and phases differing by L° (where L is a positive value less than 180); wherein a value obtained by multiplying L by M is 360, and of the M mixers, P (where P is a natural number less than M) mixers controlled by driving signals comprising pulse trains whose on-periods are adjacent are connected to the same second integration element. 
     By means of this configuration, suppression of mixer harmonic response can be achieved. 
     A multiphase mixer of the present invention, ninthly, is a multiphase mixer having the eighth configuration, having a configuration wherein each of the driving signals controlling the M mixers has an on-period ratio of 100/M (%) with respect to one cycle. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, tenthly, is a multiphase mixer having the eighth or ninth configuration, having a configuration wherein the M mixers are controlled by driving signals comprising pulse trains that are not in an on state simultaneously. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, eleventhly, is a multiphase mixer having any one of the eighth through tenth configurations, having a configuration wherein the L is 90, the M is 4, and the P is 2. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, twelfthly, is a multiphase mixer having any one of the eighth through tenth configurations, having a configuration wherein the L is 60, the M is 6, and the P is 3. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, thirteenthly, has a configuration comprising: a transconductance amplifier that converts a voltage signal to a current signal; a first integration element connected in parallel to a subsequent stage of the transconductance amplifier; and M (where M is a natural number of two or more) mixers connected in parallel with respect to the first integration element, and controlled by driving signals comprising pulse trains having the same frequency and phases differing by L° (where L is a positive value less than 180); wherein a value obtained by multiplying L by M is 360, and of the M mixers, two mixers controlled by driving signals comprising pulse trains whose on-period phase difference is 180° are connected to the same second integration element. 
     By means of this configuration, suppression of mixer harmonic response can be achieved. 
     A multiphase mixer of the present invention, fourteenthly, is a multiphase mixer having the thirteenth configuration, having a configuration wherein each of the driving signals controlling the M mixers has an on-period ratio of 100/M (%) with respect to one cycle. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
     A multiphase mixer of the present invention, fifteenthly, is a multiphase mixer having the thirteenth or fourteenth configuration, having a configuration wherein the M mixers are controlled by driving signals comprising pulse trains that are not in an on state simultaneously. 
     By means of this configuration, suppression of mixer self-mixing and suppression of mixer harmonic response can be achieved simultaneously. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1  is a configuration diagram of a conventional charge sampling circuit; 
         FIG. 2  shows an example of a multiphase mixer according to Embodiment 1; 
         FIG. 3  shows control signal waveforms of a 4-phase mixer; 
         FIG. 4  shows another example of a multiphase mixer according to Embodiment 1; 
         FIG. 5  shows an attenuation characteristic simulation result for the multiphase mixer in  FIG. 4 ; 
         FIG. 6  shows another example of a multiphase mixer according to Embodiment 1; 
         FIG. 7  shows an attenuation characteristic simulation result for the multiphase mixer in  FIG. 6 ; 
         FIG. 8  shows an example of a multiphase mixer according to Embodiment 2; 
         FIG. 9  shows control signal waveforms of a 6-phase mixer; 
         FIG. 10  shows an attenuation characteristic simulation result for the multiphase mixer in  FIG. 8 ; 
         FIG. 11  shows an example of a multiphase mixer according to Embodiment 3; 
         FIG. 12  shows an attenuation characteristic simulation result for the multiphase mixer in  FIG. 11 ; 
         FIG. 13  shows an example of a multiphase mixer according to Embodiment 4; 
         FIG. 14  shows an attenuation characteristic simulation result for the multiphase mixer in  FIG. 13 ; 
         FIG. 15  shows an example of a multiphase mixer according to Embodiment 5; 
         FIG. 16  shows an attenuation characteristic simulation result for the multiphase mixer in  FIG. 15 ; 
         FIG. 17  shows an example of a multiphase mixer according to Embodiment 6; 
         FIG. 18  shows an attenuation characteristic simulation result for the multiphase mixer in  FIG. 17 ; 
         FIG. 19  shows an example of a multiphase mixer according to Embodiment 7; 
         FIG. 20  shows an attenuation characteristic simulation result for the multiphase mixer in  FIG. 19 ; 
         FIG. 21  shows an example of a multiphase mixer according to Embodiment 8; 
         FIG. 22  shows control signal waveforms of a 4-phase mixer according to Embodiment 8; and 
         FIG. 23  shows an attenuation characteristic simulation result for the multiphase mixer in  FIG. 21 . 
     
    
    
     DESCRIPTION OF EMBODIMENTS 
     Now, embodiments of the present invention will be described in detail with reference to the accompanying drawings. 
     Embodiment 1 
     In this embodiment, a configuration is described whereby, in an orthogonal decoder configuration (4-phase mixer) that drives four charge sampling mixers using control signals with a duty of less than 50%—specifically, control signals with a duty ratio of 1/4 (25%)—an FIR filter is formed using a residual charge in a preceding stage of the charge sampling mixers, and harmonic response that is an even multiple of a local oscillation frequency is suppressed. 
       FIG. 2  shows an example of the configuration of a multiphase mixer according to this embodiment. As shown in  FIG. 2 , multiphase mixer  100  is provided with Transconductance Amplifier (TA)  101  that converts a voltage to a current, mixer  102 , mixer  103 , mixer  104 , mixer  105 , capacitance  106 , capacitance  107 , capacitance  108 , capacitance  109 , and control signal generation section  110 , performs frequency conversion of a signal input from input terminal  131 , and outputs output signal  151  from output terminal  132 , output signal  152  from output terminal  133 , output signal  153  from output terminal  134 , and output signal  154  from output terminal  135 . 
     TA  101  converts an alternating voltage input from input terminal  131  to an alternating current. Mixer  102  is connected to TA  101 , and is driven by control signal  141  output from control signal generation section  110 . Mixer  103  is connected to TA  101 , and is driven by control signal  143  output from control signal generation section  110 . Mixer  104  is connected to TA  101 , and is driven by control signal  142  output from control signal generation section  110 . Mixer  105  is connected to TA  101 , and is driven by control signal  144  output from control signal generation section  110 . Mixer  102 , mixer  103 , mixer  104 , and mixer  105  are driven by control signals, and charge capacitance  106 , capacitance  107 , capacitance  108 , and capacitance  109  with an alternating current output from TA  101  only while in an active state. That is to say, a current output from TA  101  is distributed to and charges capacitance  106 , capacitance  107 , capacitance  108 , and capacitance  109  by means of mixer  102 , mixer  103 , mixer  104 , and mixer  105  driven by control signals. Capacitance  106 , capacitance  107 , capacitance  108 , and capacitance  109  are loads corresponding to mixer  102 , mixer  103 , mixer  104 , and mixer  105 , respectively. 
     In the following description, the block comprising mixer  102 , mixer  103 , mixer  104 , and mixer  105  is referred to as the mixer block. 
       FIG. 3  shows time waveforms of control signal  141 , control signal  142 , control signal  143 , and control signal  144 . Control signal  141 , control signal  142 , control signal  143 , and control signal  144  are rectangular waves with an identical cycle having a duty ratio of 1/4, and their waveforms should preferably be shaped so that there is no Hi interval overlapping among different control signals. Below, this waveform shaping is referred to as non-overlap waveform shaping. 
     As a result of driving multiphase mixer  100  using such control signals, output signal  151 , output signal  152 , output signal  153 , and output signal  154 , corresponding to an I positive phase, I negative phase, Q positive phase, and Q negative phase, are output to output terminal  132 , output terminal  133 , output terminal  134 , and output terminal  135 , respectively. Here, since the current charging time for capacitance  106 , capacitance  107 , capacitance  108 , and capacitance  109  is 1/4 of a local oscillation signal cycle, an attenuation pole due to a current integration effect is generated at a frequency that is a constant 4 times the local oscillation signal. Also, since the multiphase mixer shown in  FIG. 2  is driven only by control signals with a duty ratio of 1/4, it has an effect of suppressing self-mixing. 
     The multiphase mixer shown in  FIG. 2  may also have the multiphase mixer configuration shown in  FIG. 4 .  FIG. 4  shows another example of the configuration of a multiphase mixer according to this embodiment. 
     Multiphase mixer  300  in  FIG. 4  differs from multiphase mixer  100  shown in  FIG. 2  in having capacitance  301  connected as an integration element in series between TA  101  and the mixer block (comprising mixers  102 ,  103 ,  104 , and  105 ). Capacitance  301  is distinct from capacitance  106 , capacitance  107 , capacitance  108 , and capacitance  109 , and below, capacitance  301  is referred to as a first integration element, and capacitance  106 , capacitance  107 , capacitance  108 , and capacitance  109  are referred to as second integration elements. By providing capacitance  301  in series between TA  101  and the mixer block, if optimal operating points differ for TA  101  and the mixer block, this can be used to set individual operating points, or to suppress 1/f noise generated by TA  101 . Other operations and functions are identical to those of multiphase mixer  100  shown in  FIG. 2 . That is to say, since the current charging time for capacitance  106 , capacitance  107 , capacitance  108 , and capacitance  109  is 1/4 of the local oscillation signal cycle, an attenuation pole due to a current integration effect is generated at a frequency that is a constant 4 times the local oscillation signal, as ringed with a solid line in  FIG. 5 . For example, in  FIG. 5 , an attenuation pole due to a current integration effect is generated at a frequency of 2.4 GHz, 4 times the local oscillation signal frequency of 600 MHz. Here,  FIG. 5  shows an attenuation characteristic simulation result for multiphase mixer  100  in  FIG. 2  or multiphase mixer  300  in  FIG. 4 . Also, since the multiphase mixer shown in  FIG. 4  is driven only by control signals with a duty ratio of 1/4, it has an effect of suppressing self-mixing. 
     A configuration will now be described, using  FIG. 6 , that makes it possible to generate an attenuation pole due to a current integration effect at a frequency that is an even multiple of a local oscillation signal, and to suppress self-mixing. 
       FIG. 6  shows another example of the configuration of a multiphase mixer according to this embodiment. Multiphase mixer  400  in  FIG. 6  differs from multiphase mixer  100  shown in  FIG. 2  in having capacitance  401  connected in series between TA  101  and mixers  102  and  103 , and having capacitance  402  connected in series between TA  101  and mixers  104  and  105 . Capacitance  401  and capacitance  402  are distinct from capacitance  106 , capacitance  107 , capacitance  108 , and capacitance  109 , and below, capacitance  401  and capacitance  402  are referred to as first integration elements, and capacitance  106 , capacitance  107 , capacitance  108 , and capacitance  109  are referred to as second integration elements. 
     When multiphase mixer  400  is mounted on a semiconductor substrate, grounding parasitic capacitances are connected to a connecting node between TA  101  and one side of capacitance  401 , a connecting node between TA  101  and one side of capacitance  402 , a connecting node between the other side of capacitance  401  and mixer  102  and mixer  103 , and a connecting node between the other side of capacitance  402  and mixer  104  and mixer  105 , respectively. 
     Here, for example, a period will be considered in which mixer  102  is active, and mixer  103 , mixer  104 , and mixer  105  are inactive. Since mixer  102  is active, charge sharing is performed between TA  101  and parasitic capacitance around mixer  102 , but by using the configuration in  FIG. 6 , charge sharing between capacitance  106  and parasitic capacitance present at a connecting node between the other side of capacitance  401  and mixer  102  and mixer  103  becomes predominant. 
     Similarly, when mixer  103  is active, charge sharing between capacitance  107  and parasitic capacitance present at a connecting node between the other side of capacitance  401  and mixer  102  and mixer  103  becomes predominant, and when mixer  104  or mixer  105  is active, charge sharing between capacitance  108  or capacitance  109  and parasitic capacitance present at a connecting node between the other side of capacitance  402  and mixer  104  and mixer  105  becomes predominant. 
     That is to say, by means of charge sharing between a load capacitance and an aforementioned predominant parasitic capacitance, an attenuation pole can be generated at a frequency that is an even multiple of the local oscillation signal apart from an attenuation pole due to a current integration effect. For example, as shown in  FIG. 7 , in addition to attenuation poles due to a current integration effect present in parts ringed with a solid line, an attenuation pole (a part ringed with a dotted line) is generated at a frequency that is an even multiple of the local oscillation signal. In the example shown in  FIG. 7 , an attenuation pole is generated at a frequency of 1.2 GHz, twice the local oscillation signal frequency of 600 MHz. Here,  FIG. 7  shows an attenuation characteristic simulation result for multiphase mixer  400  in  FIG. 6 . Also, since multiphase mixer  400  shown in  FIG. 6  is driven only by control signals with a duty ratio of 1/4, it has an effect of suppressing self-mixing. 
     In this embodiment, a configuration has been assumed in which a capacitative element is connected to a preceding stage of a plurality of mixers driven by a 180-degree phase difference, as in the case of multiphase mixer  400  shown in  FIG. 6 , with the object of generating an attenuation pole at a frequency that is an even multiple of a local oscillation signal, but if generation of an attenuation pole at a frequency that is an even multiple of a local oscillation signal is not an object, a configuration may be used in which a preceding stage of two or three mixers driven by an arbitrary phase difference is connected by a capacitance, and a filter effect due to a residual charge is arranged. 
     Also, in this embodiment, control signals with a duty ratio of 1/4 are used, but this is not a limitation, and control signals with a duty ratio of less than 1/4 may also be used. 
     Embodiment 2 
     In this embodiment, a configuration is described whereby, in a 6-phase mixer that drives six charge sampling mixers using control signals with a duty ratio of 1/6, an FIR filter is formed using a residual charge in a preceding stage of the charge sampling mixers, and harmonic response that is an even multiple of a local oscillation frequency is suppressed. 
       FIG. 8  shows an example of the configuration of a multiphase mixer according to this embodiment. As shown in  FIG. 8 , multiphase mixer  500  is provided with TA  501 , mixer  502 , mixer  503 , mixer  504 , mixer  505 , mixer  506 , mixer  507 , capacitance  508 , capacitance  509 , capacitance  510 , capacitance  511 , capacitance  512 , capacitance  513 , control signal generation section  514 , capacitance  515 , capacitance  516 , and capacitance  517 , performs frequency conversion of a signal input from input terminal  531 , and outputs output signal  551  from output terminal  532 , output signal  552  from output terminal  533 , output signal  553  from output terminal  534 , output signal  554  from output terminal  535 , output signal  555  from output terminal  536 , and output signal  556  from output terminal  537 . Below, capacitance  515 , capacitance  516 , and capacitance  517  are referred to as first integration elements, and capacitance  508 , capacitance  509 , capacitance  510 , capacitance  511 , capacitance  512 , and capacitance  512  are referred to as second integration elements. 
     TA  501  converts an alternating voltage input from input terminal  531  to an alternating current. 
     Mixer  502  is connected to TA  501 , and is driven by control signal  541  output from control signal generation section  514 . Mixer  503  is connected to TA  501 , and is driven by control signal  542  output from control signal generation section  514 . Mixer  504  is connected to TA  501 , and is driven by control signal  543  output from control signal generation section  514 . Mixer  505  is connected to TA  501 , and is driven by control signal  544  output from control signal generation section  514 . Mixer  506  is connected to TA  501 , and is driven by control signal  545  output from control signal generation section  514 . Mixer  507  is connected to TA  501 , and is driven by control signal  546  output from control signal generation section  514 . 
     Mixer  502 , mixer  503 , mixer  504 , mixer  505 , mixer  506 , and mixer  507  are driven by control signals, and charge capacitance  508 , capacitance  509 , capacitance  510 , capacitance  511 , capacitance  512 , and capacitance  513  with an alternating current output from TA  501  only while in an active state. That is to say, a current output from TA  501  is distributed to and charges capacitance  508 , capacitance  509 , capacitance  510 , capacitance  511 , capacitance  512 , and capacitance  513  by means of mixer  502 , mixer  503 , mixer  504 , mixer  505 , mixer  506 , and mixer  507  driven by control signals. That is to say, multiphase mixer  500  shown in  FIG. 8  is an extension of multiphase mixer  400  shown in  FIG. 6  from a 4-phase configuration to a 6-phase configuration. 
     Capacitance  508 , capacitance  509 , capacitance  510 , capacitance  511 , capacitance  512 , and capacitance  513  are loads corresponding to mixer  502 , mixer  503 , mixer  504 , mixer  505 , mixer  506 , and mixer  507 , respectively. 
     Capacitance  515  is a capacitance for performing capacitative coupling between TA  501  and mixer  502  and mixer  505 . Capacitance  516  is a capacitance for performing capacitative coupling between TA  501  and mixer  503  and mixer  506 . Capacitance  517  is a capacitance for performing capacitative coupling between TA  501  and mixer  504  and mixer  507 . 
       FIG. 9  shows time waveforms of control signal  541 , control signal  542 , control signal  543 , control signal  544 , control signal  545 , and control signal  546 . Control signal  541 , control signal  542 , control signal  543 , control signal  544 , control signal  545 , and control signal  546  are rectangular waves with an identical cycle having a duty ratio of 1/6, and non-overlap waveform shaping should preferably be performed on these control signals. 
     As a result of driving multiphase mixer  500  using such control signals, output signal  551 , output signal  552 , output signal  553 , output signal  554 , output signal  555 , and output signal  556 , having a  60 -degree phase difference relationship in a post-frequency-conversion baseband, are output to output terminal  532 , output terminal  533 , output terminal  534 , output terminal  535 , output terminal  536 , and output terminal  537 , respectively. 
     Here, since the current charging time for capacitance  508 , capacitance  509 , capacitance  510 , capacitance  511 , capacitance  512 , and capacitance  513  is 1/6 of a local oscillation signal cycle, an attenuation pole due to a current integration effect is generated at a frequency that is a constant 6 times the local oscillation signal. For example, an attenuation pole due to a current integration effect is generated at a frequency of 3.6 GHz, 6 times the local oscillation signal frequency of 600 MHz, as ringed with a solid line in  FIG. 10 . Here,  FIG. 10  shows an attenuation characteristic simulation result for multiphase mixer  500  in  FIG. 8 . 
     Capacitance  515 , capacitance  516 , and capacitance  517 , which are first integration elements, selectively use a charge accumulated in a parasitic capacitance that performs charge sharing with a load capacitance while a mixer is activated by coupling mixers driven by control signals with a 180-degree phase difference on one side of a capacitance from among mixers driven by control signals with different phases. That is to say, as described in Embodiment 1, an attenuation pole can be generated at a frequency that is an even multiple of the local oscillation signal apart from an attenuation pole due to a current integration effect. For example, as shown in  FIG. 10 , in addition to an attenuation pole due to a current integration effect present in a part ringed with a solid line, an attenuation pole (a part ringed with a dotted line) is generated at a frequency that is an even multiple of the local oscillation signal. In the example shown in  FIG. 10 , an attenuation pole is generated at a frequency of 1.2 GHz, twice the local oscillation signal frequency of 600 MHz. Also, since the multiphase mixer shown in  FIG. 8  is driven only by control signals with a duty ratio of 1/6, it has an effect of suppressing self-mixing. 
     In this embodiment, a configuration has been assumed in which a capacitative element is connected to a preceding stage of a plurality of mixers driven by a 180-degree phase difference, as in the case of multiphase mixer  500  shown in  FIG. 8 , with the object of generating an attenuation pole at a frequency that is an even multiple of a local oscillation signal, but if generation of an attenuation pole at a frequency that is an even multiple of a local oscillation signal is not an object, a configuration may be used in which a preceding stage of a plurality of mixers driven by an arbitrary phase difference is connected by a capacitance, and a filter effect due to a residual charge is arranged. 
     Also, in this embodiment, control signals with a duty ratio of 1/6 are used, but this is not a limitation, and control signals with a duty ratio of less than 1/6 may also be used. 
     Here, a 4-phase mixer configuration has been described in Embodiment 1, and a 6-phase mixer configuration in this embodiment, but an attenuation pole can be generated at a frequency that is an even multiple of the local oscillation signal, apart from an attenuation pole due to a current integration effect, by coupling preceding stages of mixers activated by a 180-degree phase difference, by a capacitance functioning as a first integration element, in an even-numbered-phase mixer configuration. 
     Embodiment 3 
     In this embodiment, a configuration is described whereby, in a 4-phase mixer driven using control signals with a duty ratio of less than 50%—specifically, control signals with a duty ratio of 1/4—harmonic response that is an even multiple of a local oscillation frequency is suppressed by charging the same load capacitance with outputs of mixers activated by adjacent phase differences. 
       FIG. 11  shows an example of the configuration of a multiphase mixer according to this embodiment. As shown in  FIG. 11 , multiphase mixer  700  is provided with TA  101 , mixer  102 , mixer  103 , mixer  104 , and mixer  105 , capacitance  301 , capacitance  701 , capacitance  702 , and control signal generation section  110 , performs frequency conversion of a signal input from input terminal  131 , and outputs output signal  751  from output terminal  732 , and output signal  752  from output terminal  733 . Below, capacitance  301  is referred to as a first integration element, and capacitance  701  and capacitance  702  are referred to as second integration elements. 
     Here, the operation and function of configuration elements assigned the same reference numbers as in  FIG. 2  of Embodiment 1 or  FIG. 4  of Embodiment 2 are identical to those in  FIG. 2  or  FIG. 4 , and descriptions thereof are omitted. 
     In the following description, the block comprising mixer  102 , mixer  103 , mixer  104 , and mixer  105  is referred to as the mixer block. Here, capacitance  701  is a load corresponding to mixer  102  and mixer  104 , and capacitance  702  is a load corresponding to mixer  103  and mixer  104 . 
     As a result of driving multiphase mixer  700  using the control signals shown in  FIG. 3 , output signal  751  and output signal  752  having a 180-degree phase difference relationship in a post-frequency-conversion baseband are output to output terminal  732  and output terminal  733 , respectively. 
     At this time, mixer  102  and mixer  104  connected to capacitance  701  are controlled by driving signals comprising pulse trains with adjacent on-periods. Also, mixer  103  and mixer  105  connected to capacitance  702  are controlled by driving signals comprising pulse trains with adjacent on-periods. 
     Although it is desirable for non-overlap waveform shaping to be performed for control signal  141  and control signal  144 , and control signal  142  and control signal  143 , it is not absolutely necessary for non-overlap waveform shaping to be performed for control signal  141  and control signal  142 , and control signal  143  and control signal  144 . 
     By means of this kind of configuration, charging is performed successively via mixers driven by adjacent control signals, and therefore the current charging time for capacitance  701  and capacitance  702  is ½ of a local oscillation signal cycle, and an attenuation pole due to a current integration effect is generated at a frequency that is an even multiple of the local oscillation signal frequency. For example, in  FIG. 12 , an attenuation pole due to a current integration effect is generated at a frequency of 1.2 GHz, twice the local oscillation signal frequency of 600 MHz. Here,  FIG. 12  shows an attenuation characteristic simulation result for multiphase mixer  700  in  FIG. 11 . Also, since the multiphase mixer shown in  FIG. 11  is driven only by control signals with a duty ratio of 1/4, it has an effect of suppressing self-mixing. 
     In this embodiment, control signals with a duty ratio of 1/4 are used, but this is not a limitation, and control signals with a duty ratio of less than 1/4 may also be used. 
     Also, in this embodiment, if optimal operating points differ for TA  101  and the mixer block, capacitance  301  is used to set individual operating points, or to suppress 1/f noise generated by TA  101 , but this is not a limitation, and capacitance  301  may be eliminated, and TA  101  and the mixer block may be connected without the intermediation of a capacitance. 
     Embodiment 4 
     In this embodiment, a configuration is described whereby, in a 6-phase mixer driven using control signals with a duty ratio of less than 50%—specifically, control signals with a duty ratio of 1/6—harmonic response that is an even multiple of a local oscillation frequency is suppressed by charging the same load capacitance with outputs of mixers activated by adjacent phase differences. That is to say, a configuration is shown in which 4-phase mixer  700  shown in Embodiment 3 is extended to a 6-phase mixer. 
       FIG. 13  shows an example of the configuration of a multiphase mixer according to this embodiment. As shown in  FIG. 13 , multiphase mixer  800  is provided with TA  501 , mixer  502 , mixer  503 , mixer  504 , mixer  505 , mixer  506 , mixer  507 , capacitance  801 , capacitance  802 , control signal generation section  514 , and capacitance  803 , performs frequency conversion of a signal input from input terminal  531 , and outputs output signal  851  from output terminal  832 , and output signal  852  from output terminal  833 . Here, the operation and function of configuration elements assigned the same reference numbers as in  FIG. 8  of Embodiment 2 are identical to those in  FIG. 8 , and descriptions thereof are omitted. 
     Also, in the following description, the block comprising mixer  502 , mixer  503 , mixer  504 , mixer  505 , mixer  506 , and mixer  507  is referred to as the mixer block. 
     Capacitance  801  is a load corresponding to mixer  502 , mixer  503 , and mixer  504 , and capacitance  802  is a load corresponding to mixer  505 , mixer  506 , and mixer  507 . Capacitance  803  is a capacitive coupling capacitance that is connected in series between TA  501  and the mixer block, and is used for operating point optimization for the various circuits, or to suppress 1/f noise generated by TA  501 . 
     As a result of driving multiphase mixer  800  using the control signals shown in  FIG. 9 , output signal  851  and output signal  852  having a  180 -degree phase difference relationship in a post-frequency-conversion baseband are output to output terminal  832  and output terminal  833 , respectively. 
     Although it is desirable for non-overlap waveform shaping to be performed for control signal  541  and control signal  546 , and control signal  543  and control signal  544 , it is not absolutely necessary for non-overlap waveform shaping to be performed for other adjacent control signals. 
     By means of this kind of configuration, charging is performed successively via mixers driven by adjacent control signals, and therefore the current charging time for capacitance  801  and capacitance  802  is ½ of a local oscillation signal cycle, and an attenuation pole due to a current integration effect is generated at a frequency that is an even multiple of the local oscillation signal frequency. For example, an attenuation pole due to a current integration effect is generated at a frequency that is an even multiple of the local oscillation signal frequency, as ringed with a solid line in  FIG. 14 . For example, in  FIG. 14 , an attenuation pole due to a current integration effect is generated at a frequency of 1.2 GHz, twice the local oscillation signal frequency of 600 MHz. Here,  FIG. 14  shows an attenuation characteristic simulation result for multiphase mixer  800  in  FIG. 13 . Also, since the multiphase mixer shown in  FIG. 13  is driven only by control signals with a duty ratio of 1/6, it has an effect of suppressing self-mixing. 
     In this embodiment, control signals with a duty ratio of 1/6 are used, but this is not a limitation, and control signals with a duty ratio of less than 1/6 may also be used. 
     Also, in this embodiment, capacitance  803  is used to perform capacitive coupling between TA  501  and the mixer block, but this is not a limitation, and capacitance  803  may be eliminated, and TA  501  and the mixer block may be connected without the intermediation of a capacitance. 
     Embodiment 5 
     In this embodiment, a configuration is described whereby, in a 6-phase mixer driven using control signals with a duty ratio of less than 50%—specifically, control signals with a duty ratio of 1/6—harmonic response that is an integer 3 multiple of a local oscillation frequency is suppressed by charging the same load capacitance with outputs of mixers activated by a 180-degree phase difference. 
       FIG. 15  shows an example of the configuration of a multiphase mixer according to this embodiment. As shown in  FIG. 15 , multiphase mixer  900  is provided with TA  501 , mixer  502 , mixer  503 , mixer  504 , mixer  505 , mixer  506 , mixer  507 , capacitance  901 , capacitance  902 , capacitance  903 , control signal generation section  514 , and capacitance  803 , performs frequency conversion of a signal input from input terminal  531 , and outputs output signal  951  from output terminal  932 , output signal  952  from output terminal  933 , and output signal  953  from output terminal  934 . Here, the operation and function of configuration elements assigned the same reference numbers as in  FIG. 13  of Embodiment 4 are identical to those in  FIG. 13 , and descriptions thereof are omitted. 
     Also, in the following description, the block comprising mixer  502 , mixer  503 , mixer  504 , mixer  505 , mixer  506 , and mixer  507  is referred to as the mixer block. 
     Capacitance  901  is a load corresponding to mixer  502  and mixer  505 , capacitance  902  is a load corresponding to mixer  503  and mixer  506 , and capacitance  903  is a load corresponding to mixer  504  and mixer  507 . 
     Here, it is desirable for inter-control-signal non-overlap waveform shaping to be performed for the control signals shown in  FIG. 9 . By means of this kind of configuration, current charging is performed via mixers. The charging time is ⅓ of a local oscillation signal cycle, and an attenuation pole due to a current integration effect is generated at a frequency that is an even multiple of the local oscillation signal frequency. For example, an attenuation pole due to a current integration effect is generated at a frequency that is a constant 3 multiple of the local oscillation signal frequency, as ringed with a solid line in  FIG. 16 . For example, in  FIG. 16 , an attenuation pole due to a current integration effect is generated at a frequency of 1.8 GHz, three times the local oscillation signal frequency of 600 MHz. Here,  FIG. 16  shows an attenuation characteristic simulation result for multiphase mixer  900  in  FIG. 15 . Also, since the multiphase mixer shown in  FIG. 15  is driven only by control signals with a duty ratio of 1/6, it has an effect of suppressing self-mixing. 
     In this embodiment, control signals with a duty ratio of 1/6 are used, but this is not a limitation, and control signals with a duty ratio of less than 1/6 may also be used. 
     Also, in this embodiment, capacitance  803  is used to perform capacitive coupling between TA  501  and the mixer block, but this is not a limitation, and capacitance  803  may be eliminated, and TA  501  and the mixer block may be connected without the intermediation of a capacitance. 
     Furthermore, in Embodiment 4, a configuration is used in which the outputs of three mixers driven by 60-degree phase differences are connected, as in multiphase mixer  800  shown in  FIG. 13 , with the object of generating an attenuation pole at a frequency that is an even multiple of a local oscillation signal. In contrast, in this embodiment, a configuration is used in which the outputs of two mixers driven by a 180-degree phase difference are connected, as in multiphase mixer  900  shown in  FIG. 15 , with the object of generating an attenuation pole at a frequency that is a constant 3 multiple of a local oscillation signal. However, if generation of an attenuation pole at a frequency that is an even multiple, or a constant 3 multiple, of a local oscillation signal is not an object, for instance, a configuration may be used whereby load capacitance charging is performed with the outputs of a plurality of mixers driven by arbitrary phase differences connected. 
     Embodiment 6 
     This embodiment is a combination of Embodiment 2 and Embodiment 4. 
       FIG. 17  shows an example of the configuration of a multiphase mixer according to this embodiment. As shown in  FIG. 17 , multiphase mixer  1000  is provided with TA  501 , mixer  502 , mixer  503 , mixer  504 , mixer  505 , mixer  506 , mixer  507 , capacitance  801 , capacitance  802 , control signal generation section  514 , capacitance  1001 , capacitance  1002 , and capacitance  1003 , performs frequency conversion of a signal input from input terminal  531 , and outputs output signal  851  from output terminal  832 , and output signal  852  from output terminal  833 . Here, the operation and function of configuration elements assigned the same reference numbers as in  FIG. 13  of Embodiment 4 are identical to those in  FIG. 13 , and descriptions thereof are omitted. 
     Capacitance  1001  is a capacitance for performing capacitive coupling between TA  501  and mixer  502  and mixer  505 . Capacitance  1002  is a capacitance for performing capacitive coupling between TA  501  and mixer  503  and mixer  506 . 
     Capacitance  1003  is a capacitance for performing capacitive coupling between TA  501  and mixer  504  and mixer  507 . Below, capacitance  1001 , capacitance  1002 , and capacitance  1003  are referred to as first integration elements, and capacitance  801  and capacitance  802  are referred to as second integration elements. 
     As a result of driving multiphase mixer  1000  using the control signals shown in  FIG. 9 , output signal  851  and output signal  852  having a 180-degree phase difference relationship in a post-frequency-conversion baseband are output to output terminal  832  and output terminal  833 , respectively. 
     At this time, mixer  502 , mixer  503 , and mixer  504  connected to capacitance  801  are controlled by driving signals comprising pulse trains with adjacent on-periods. Also, mixer  505 , mixer  506 , and mixer  507  connected to capacitance  802  are controlled by driving signals comprising pulse trains with adjacent on-periods. 
     As the current charging time for capacitance  801  and capacitance  802  is ½ of a local oscillation signal cycle, an attenuation pole due to a current integration effect is generated at a frequency that is an even multiple of the local oscillation signal frequency. 
     Also, capacitance  1001 , capacitance  1002 , and capacitance  1003  selectively use a charge accumulated in a parasitic capacitance that performs charge sharing with a load capacitance while a mixer is activated by coupling mixers driven by control signals with a 180-degree phase difference on one side of a capacitance from among mixers driven by control signals with different phases. That is to say, as described in Embodiment 2, an attenuation pole can be generated at a frequency that is an even multiple of the local oscillation signal in addition to an attenuation pole due to a current integration effect, and attenuation can be improved. For example, an attenuation pole due to a current integration effect is generated at a frequency that is an even multiple of the local oscillation signal, as ringed with a solid line in  FIG. 18 . For example, in  FIG. 18 , an attenuation pole due to a current integration effect is generated at a frequency of 1.2 GHz, twice the local oscillation signal frequency of 600 MHz. Here,  FIG. 18  shows an attenuation characteristic simulation result for multiphase mixer  1000  in  FIG. 17 . Also, since the multiphase mixer shown in  FIG. 17  is driven only by control signals with a duty ratio of 1/6, it has an effect of suppressing self-mixing. 
     In this embodiment, control signals with a duty ratio of 1/6 are used, but this is not a limitation, and control signals with a duty ratio of less than 1/6 may also be used. 
     Embodiment 7 
     This embodiment is a combination of Embodiment 2 and Embodiment 5. 
       FIG. 19  shows an example of the configuration of a multiphase mixer according to this embodiment. As shown in  FIG. 19 , multiphase mixer  1100  is provided with TA  501 , mixer  502 , mixer  503 , mixer  504 , mixer  505 , mixer  506 , mixer  507 , capacitance  901 , capacitance  902 , capacitance  903 , control signal generation section  514 , capacitance  1101 , capacitance  1102 , and capacitance  1103 , performs frequency conversion of a signal input from input terminal  531 , and outputs output signal  951  from output terminal  932 , output signal  952  from output terminal  933 , and output signal  953  from output terminal  934 . Here, the operation and function of configuration elements assigned the same reference numbers as in  FIG. 15  of Embodiment 5 are identical to those in  FIG. 15 , and descriptions thereof are omitted. 
     Capacitance  1101  is a capacitance for performing capacitive coupling between TA  501  and mixer  502  and mixer  505 . Capacitance  1102  is a capacitance for performing capacitive coupling between TA  501  and mixer  503  and mixer  506 . Capacitance  1103  is a capacitance for performing capacitive coupling between TA  501  and mixer  504  and mixer  507 . The current charging time for capacitance  901  and capacitance  902  is ⅓ of a local oscillation signal cycle, and therefore an attenuation pole due to a current integration effect is generated at a frequency that is a constant 3 multiple of the local oscillation signal frequency. Below, capacitance  1101 , capacitance  1102 , and capacitance  1103  are referred to as first integration elements, and capacitance  901 , capacitance  902 , and capacitance  903  are referred to as second integration elements. 
     As a result of driving multiphase mixer  1100  using the control signals shown in  FIG. 9 , mixer  502  and mixer  505  connected to capacitance  901  are controlled by driving signals comprising pulse trains with an on-period phase difference of 180°, mixer  503  and mixer  506  connected to capacitance  902  are controlled by driving signals comprising pulse trains with an on-period phase difference of 180°, and mixer  505  and mixer  507  connected to capacitance  903  are controlled by driving signals comprising pulse trains with an on-period phase difference of 180°. 
     Also, capacitance  1101 , capacitance  1102 , and capacitance  1103  selectively use a charge accumulated in a parasitic capacitance that performs charge sharing with a load capacitance while a mixer is activated by coupling mixers driven by control signals with a 180-degree phase difference on one side of a capacitance from among mixers driven by control signals with different phases. That is to say, as described in Embodiment 2, an attenuation pole can be generated at a frequency that is an even multiple of the local oscillation signal in addition to an attenuation pole due to a current integration effect. For example, as shown in  FIG. 20 , in addition to an attenuation pole due to a current integration effect present in a part ringed with a solid line, an attenuation pole (a part ringed with a dotted line) is generated at a frequency that is an even multiple of the local oscillation signal. In the example shown in  FIG. 20 , an attenuation pole is generated at a frequency of 1.2 GHz, twice the local oscillation signal frequency of 600 MHz. Here,  FIG. 20  shows an attenuation characteristic simulation result for multiphase mixer  1100  in  FIG. 19 . Also, since the multiphase mixer shown in  FIG. 19  is driven only by control signals with a duty ratio of 1/6, it has an effect of suppressing self-mixing. 
     In this embodiment, control signals with a duty ratio of 1/6 are used, but this is not a limitation, and control signals with a duty ratio of less than 1/6 may also be used. 
     Embodiment 8 
     In this embodiment, a configuration is described whereby, in a 4-phase mixer that is driven using control signals with a duty ratio of 25%, frequency conversion is executed based on a frequency that is twice the control signal frequency by charging the same load capacitance with outputs of mixers activated by a 180-degree phase difference. 
       FIG. 21  shows an example of the configuration of a multiphase mixer according to this embodiment. 
     As shown in  FIG. 21 , multiphase mixer  1200  is provided with TA  1201 , mixer  1202 , mixer  1203 , mixer  1204 , mixer  1205 , capacitance  1206 , capacitance  1207 , control signal generation section  1208 , and capacitance  1209 , performs frequency conversion of a signal input from input terminal  1231 , and outputs output signal  1251  from output terminal  1232 , and output signal  1252  from output terminal  1233 . Below, capacitance  1209  is referred to as a first integration element, and capacitance  1206  and capacitance  1207  are referred to as second integration elements. 
     TA  1201  converts an alternating voltage input from input terminal  1231  to an alternating current. Mixer  1202  is connected to TA  1201 , and is driven by control signal  1241  output from control signal generation section  1208 . Mixer  1203  is connected to TA  1201 , and is driven by control signal  1243  output from control signal generation section  1208 . Mixer  1204  is connected to TA  1201 , and is driven by control signal  1242  output from control signal generation section  1208 . Mixer  1205  is connected to TA  1201 , and is driven by control signal  1244  output from control signal generation section  1208 . 
     Mixer  1202 , mixer  1203 , mixer  1204 , and mixer  1205  are driven by control signals, and charge capacitance  1206  and capacitance  1207  with an alternating current output from TA  1201  only while in an active state. That is to say, a current output from TA  1201  is distributed to and charges capacitance  1206  and capacitance  1207  by means of mixer  1202 , mixer  1203 , mixer  1204 , and mixer  1205  driven by control signals. 
     Below, the block comprising mixer  1202 , mixer  1203 , mixer  1204 , and mixer  1205  is referred to as the mixer block. 
     Capacitance  1206  is a load corresponding to mixer  1202  and mixer  1203 , and capacitance  1207  is a load corresponding to mixer  1204  and mixer  1205 . 
     Capacitance  1209  is a capacitance for performing capacitive coupling between TA  1201  and mixer  1202 , mixer  1203 , mixer  1204 , and mixer  1205 . 
       FIG. 22  shows time waveforms of control signal  1241 , control signal  1242 , control signal  1243 , and control signal  1244 . Control signal  1241 , control signal  1242 , control signal  1243 , and control signal  1244  are rectangular waves with an identical cycle having a duty ratio of 1/4, and their waveforms should preferably be shaped so that there is no Hi interval overlapping among different control signals (that is, subjected to non-overlap waveform shaping). As a result of driving multiphase mixer  1200  using such control signals, output signal  1251  and output signal  1252  having a 180-degree phase difference relationship in a post-frequency-conversion baseband are output to output terminal  1232  and output terminal  1233  respectively. 
     Here, the current charging frequency for capacitance  1206  and capacitance  1207  is a multiple of the control signal cycle, and frequency conversion is executed at a frequency that is a multiple of the control signal frequency. For example,  FIG. 23  shows an attenuation characteristic simulation result for multiphase mixer  1200  in  FIG. 21  when the control signal frequency is 300 MHz (control signal cycle: 1/300 MHz). As shown in  FIG. 23 , a fundamental response is output at a frequency that is a multiple of the control signal frequency. 
     In this embodiment, control signals with a duty ratio of 1/4 are used, but this is not a limitation, and control signals with a duty ratio of less than 1/4 may also be used. 
     Also, in this embodiment, capacitance  1209  is used to perform capacitive coupling between TA  1201  and the mixer block, but this is not a limitation, and TA  1201  and the mixer block may be directly connected without the intermediation of capacitive coupling. 
     In Embodiment 1 through Embodiment 8, examples have been shown in which NMOS switches are used as mixers, but a configuration employing PMOS switches or CMOS switches using PMOS and NMOS in a complementary fashion may also implemented. Connections may also be made with the source terminal and drain terminal reversed. With NMOS switches being used, the description has assumed that a mixer is placed in an active state while a control signal is Hi (in the high period of a rectangular pulse), but if a PMOS configuration or a CMOS configuration with PMOS and NMOS utilized in a complementary fashion is used for a mixer, it goes without saying that a mixer using PMOS can be placed in an active state by reading “Hi period” as “Low period” (the low period of a rectangular pulse) in the description. In any case, a period during which a mixer is in an active state can be referred to as an on-period, and a period during which a mixer is in an inactive state can be referred to as an off-period. 
     Also, in Embodiment 1 through Embodiment 8, it has been assumed that a TA input signal is a single-phase input, but this is not a limitation, and provision may also be made for a differential input signal to be utilized using a differential-configuration TA, and for mixers to be arranged to suit a differential input signal and for a TA input signal to be made a differential input. 
     Furthermore, a direct sampling circuit may be configured by connecting a sampling circuit shown in Non-Patent Literature 7, for example, to a subsequent stage of a multiphase mixer shown in Embodiment 1 through Embodiment 8. 
     The disclosure of Japanese Patent Application No. 2009-017899, filed on Jan. 29, 2009, including the specification, drawings and abstract, is incorporated herein by reference in its entirety. 
     INDUSTRIAL APPLICABILITY 
     A multiphase mixer of the present invention is suitable for use in a self-mixing suppression technology for a mixer used in a high frequency processing section in a radio communication system, a mixer harmonic response suppression technology, and so forth. 
     REFERENCE SIGNS LIST 
     
         
           100 ,  300 ,  400 ,  500 ,  700 ,  800 ,  900 ,  1000 ,  1100 ,  1200  Multiphase mixer 
           101 ,  501 ,  1201  TA 
           102 ,  103 ,  104 ,  105 ,  502 ,  503 ,  504 ,  505 ,  506 ,  507 ,  1202 ,  1203 ,  1204 ,  1205  Mixer 
           106 ,  107 ,  108 ,  109 ,  301 ,  401 ,  402 ,  508 ,  509 ,  510 ,  511 ,  512 ,  513 ,  515 ,  516 ,  517 ,  701 ,  702 ,  801 ,  802 ,  803 ,  901 ,  902 ,  903 ,  1001 ,  1002 ,  1003 ,  1101 ,  1102 ,  1103 ,  1206 ,  1207 ,  1209  Capacitance 
           110 ,  514 ,  1208  Control signal generation section 
           131 ,  531 ,  1231  Input terminal 
           132 ,  133 ,  134 ,  135 ,  532 ,  533 ,  534 ,  535 ,  536 ,  537 ,  732 ,  733 ,  832 ,  833 ,  932 ,  933 ,  934 ,  1232 ,  1233  Output terminal 
           141 ,  142 ,  143 ,  144 ,  541 ,  542 ,  543 ,  544 ,  545 ,  546 ,  1241 ,  1242 ,  1243 ,  1244  Control signal 
           151 ,  152 ,  153 ,  154 ,  551 ,  552 ,  553 ,  554 ,  555 ,  556 ,  751 ,  752 ,  851 ,  852 ,  951 ,  952 ,  953 ,  1251 ,  1252  Output signal