Abstract:
A switching power supply device has a drive circuit that can minimize a loss while being compact. The drive circuit turns on and off a high-side switching element (MOSFET) according to a positive or negative voltage developed at a tertiary winding of a transformer. The drive circuit includes a control unit that detects the development time, during which the negative voltage is developed at the tertiary winding, as the on time of a low-side switching element, and makes the on time of the high-side switching element nearly or substantially equal to the development time.

Description:
BACKGROUND 
       FIG. 14  shows an embodiment of a DC-DC converter similar to the one disclosed, for example, in JP-A-2001-224171 (pages 7 and 8 and FIG. 1). As illustrated, a series circuit, including a metal oxide semiconductor field-effect transistor (MOSFET)  1  and a MOSFET  2 , is connected in parallel with a direct current (DC) power supply  3 . A series circuit, including a capacitor  4 , an inductor  5 , and a primary winding P 1  of a transformer  6 , is connected in parallel with the MOSFET  2 . Diodes  7 ,  8  and a capacitor  9  that rectify and smooth a positive or negative voltage developed at secondary windings S 1  and S 2  of a transformer  6  are connected to the secondary windings. A series circuit, including the capacitor  4 , inductor  5 , and primary winding P 1  of the transformer  6  is connected in parallel with the MOSFET  1 . The inductor  5 , which is connected in series with the primary winding P 1 , can be substituted with the leakage inductance of the transformer  6 . 
     A feedback signal is transferred to a control circuit  11  via a voltage detection circuit  10  to keep a voltage V 0  across the capacitor  9  constant. The control circuit  11  includes a comparison circuit that compares the feedback signal with a predetermined value, and a pulse generator that generates a gate pulse to be fed to the MOSFET  1  based on the results of the comparison. 
     In  FIG. 14 , the control circuit  11  and a drive circuit  12  alternately turns on and off the MOSFETs  1  and  2  at a duty cycle of 50%. An output voltage is regulated by varying the driving frequency for the MOSFETs  1  and  2 . An integrated circuit (IC) capable of withstanding a high voltage and having a level shift capability can be used as a means for transferring a signal, with which the MOSFET  2  on a high-potential side (hereinafter high-side) is turned on or off, from the control circuit  11  to the drive circuit  12 . However, since the IC capable of withstanding a high voltage is relatively expensive, the use of the IC increases the overall cost of a switching power supply device. 
       FIG. 15  shows an embodiment similar to the one disclosed, for example, in JP-A-2002-209381 (page 4 and FIG. 1) (refer to column 4, lines 10-23, and FIG. 1 of corresponding U.S. Pat. No. 6,483,722), that can reduce the cost. In the illustrated embodiment, the transformer  6  has a tertiary winding P 2 , and the high-side MOSFET  2  is driven via the drive circuit  12  according to a positive or negative voltage developed at the tertiary winding P 2 . In JP-A-2002-209381, the drive circuit  12  includes a resistor. The low-potential side (hereinafter low-side) MOSFET  1  is driven so that it is turned on after the MOSFET  2  is turned off. Thus, the two MOSFETs are alternately turned on and off. Owing to the foregoing configuration, the control circuit  11  can employ an inexpensive IC capable of withstanding a low voltage. Consequently, the cost of the switching power supply device can be reduced. 
     In the circuitry shown in  FIG. 15 , the on time of the MOSFET  2  is determined with a voltage developed at the tertiary winding P 2  of the transformer  6 . The on times of the MOSFET  1  and MOSFET  2  are usually different from each other. Consequently, current flowing through the MOSFETs  1  and  2  and current flowing through the rectification diodes  7  and  8  are unbalanced. This, however, undesirably increases the losses occurring in the MOSFETs and rectification diodes. 
     In particular, when a main AC voltage is rectified and smoothed as a DC supply voltage to be fed from the DC power supply  3 , since the AC voltage varies within a certain range, the ratio of the maximum value of the DC supply voltage to the minimum value thereof gets almost doubled. The higher the DC supply voltage is, the more remarkable the unbalance between the currents flowing through the MOSFETs and rectification diodes becomes. 
     Accordingly, there remains a need for an improved switching power supply device, or more particularly, to a technology for designing a half-bridge direct current-to-direct current (DC-DC) converter, that can resolve the unbalance between the currents flowing through MOSFETs and the currents flowing through rectification diodes so as to thus minimize losses, as well as being compact and low-cost. The present disclosure addresses this need. 
     SUMMARY OF THE INVENTION 
     The present invention relates to a switching power supply device and a drive circuit thereof. 
     One aspect of the present invention is the switching power supply device. The device can include a first switching element adapted to be connected to a negative electrode of a DC power supply, a second switching element connected to the first switching element and adapted to be connected to a positive electrode of the DC power supply, a transformer including a primary winding, a secondary winding, and a tertiary winding, and a series circuit, including a capacitor, an inductor, and the primary winding connected in series, connected in parallel with one of the first switching element or the second switching element, adapted to rectify and smooth a positive or negative voltage developed at the secondary winding by alternately turning on and off the first switching element and second switching element and provide a DC output. The device further includes a drive circuit adapted to turn on or off the second switching element according to a positive or negative voltage developed at the tertiary winding. The drive circuit includes a control unit adapted to detect a development time, during which the negative voltage is developed at the tertiary winding, as the on time of the first switching element, and make the on time of the second switching element become substantially equal to the on time of the first switching element. 
     The control unit can include at least one constant current circuit and a timer capacitor, charges or discharges the timer capacitor according to the positive or negative voltage developed at the tertiary winding, and controls the turning off timing of the second switching element based on the voltage at the terminals of the timer capacitor. The control unit can include a setting means for setting the minimum value of the on time of the high-side switching element. 
     Another aspect of the present invention is the drive circuit for a switching power supply device having a half-bridge configuration. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWING 
         FIG. 1  is a circuit diagram showing an embodiment according to the present invention. 
         FIG. 2  is a circuit diagram showing an embodiment of a control unit shown in  FIG. 1 . 
         FIG. 3  is a circuit diagram showing an embodiment of a charge/discharge circuit shown in  FIG. 2 . 
         FIG. 4  is a circuit diagram showing another embodiment of the charge/discharge circuit shown in  FIG. 2 . 
         FIG. 5  is a circuit diagram showing an embodiment of a constant current circuit shown in  FIG. 3  and  FIG. 4 . 
         FIG. 6  is a circuit diagram showing another embodiment of the constant current circuit shown in  FIG. 3  and  FIG. 4 . 
         FIG. 7  is a circuit diagram showing still another embodiment of the constant current circuit shown in  FIG. 3  and  FIG. 4 . 
         FIG. 8  shows waveforms for use in explaining actions to be performed in the embodiment shown in  FIG. 1 . 
         FIG. 9  is a circuit diagram showing another embodiment of the control unit. 
         FIG. 10  shows waveforms for use in explaining actions (part  1 ) to be performed in the control unit shown in  FIG. 9 . 
         FIG. 11  shows waveforms for use in explaining actions (part  2 ) to be performed in the control unit shown in  FIG. 9 . 
         FIG. 12  is a circuit diagram showing an embodiment of a charge/discharge circuit shown in  FIG. 9 . 
         FIG. 13  is a circuit diagram showing another embodiment of the charge/discharge circuit shown in  FIG. 9 . 
         FIG. 14  is a circuit diagram showing the first related art. 
         FIG. 15  is a circuit diagram showing the second related art. 
     
    
    
     DETAILED DESCRIPTION 
       FIG. 1 , which is a circuit diagram showing an embodiment according to the present invention, illustrates a major portion of a switching power supply device. Specifically, the power supply device according to the present invention incorporates a half-bridge direct current-to-direct current (DC-DC) converter and can resolve the unbalance between the currents flowing through MOSFETs and the currents flowing through rectification diodes to minimize losses, while making the device compact and economical. 
     The switching power supply device according to the present invention can incorporate the device illustrated in  FIG. 15 , with a drive circuit  12   a  according to the present replacing or substituting the drive circuit  12 . The drive circuit  12   a  can include resistors  101 ,  105 ,  106 , diodes  102 ,  104 , a PNP transistor  103 , and a control unit  100 . When a voltage VP 2  developed at a tertiary winding P 2  of the transformer  6  exceeds a gate threshold for the MOSFET  2 , the drive circuit  12   a  turns on the MOSFET  2 . When the voltage VP 2  falls below the gate threshold for the MOSFET  2 , the drive circuit  12   a  turns off the MOSFET  2 . 
     The control unit  100  controls the turning off timing of the MOSFET  2 . The PNP transistor  103  is turned on at a timing designated by the control unit  100 . Thus, the charge stored due to the gate input capacitance of the MOSFET  2  can be quickly discharged to turn off the MOSFET  2 . Specifically, whichever of the timing when the voltage VP 2  developed at the tertiary winding P 2  falls below the gate threshold for the MOSFET  2 , or the timing designated by the control unit  100  comes earlier, the MOSFET  2  is turned off at the earlier timing. 
       FIG. 2  shows a specific embodiment  100   a  of the control unit  100 . The control unit  100   a  includes a capacitor  201 , a diode  202 , a resistor  203 , an NPN transistor  204 , and a charge/discharge circuit  200  that charges or discharges the capacitor  201 . Charging or discharging is selected based on the polarity of the voltage between terminals a and b. The capacitor  201  is a timer capacitor that is charged or discharged by the charge/discharge circuit. 
       FIG. 3  illustrates an embodiment  200   a  of the charge/discharge circuit  200 . The charge/discharge circuit  200   a  includes constant current circuits  400   a ,  400   b  having the circuitry shown in any of  FIG. 5  to  FIG. 7 , and diodes  401 ,  402 . The embodiment of the constant current circuits  400   a ,  400   b  shown in  FIG. 5  includes a constant current diode  501 . The embodiment shown in  FIG. 6  includes an N-channel junction field-effect transistor (JFET)  502  and a resistor  503  (the resistance of the resistor  503  can be zero). The embodiment shown in  FIG. 7  includes NPN transistors  504 ,  505  and resistors  506 ,  507 . 
       FIG. 8  illustrates actions to be performed in the embodiment shown in  FIG. 1 , referring to a gate voltage VGS (Q 1 ) of the MOSFET  1 , a drain-source voltage VQ 1  of the MOSFET  1 , a voltage VP 2  at the tertiary winding of the transformer  6 , a voltage V 1  at the terminals of the capacitor  201 , and a collector-emitter voltage Voff of the NPN transistor  204 . During the on period T 1  of the MOSFET  1 , a pulse width is controlled (PWM control) so that the output voltage Vo of a DC-DC converter will remain constant. During the on period T 1 , the voltage VP 2  is negative, and the capacitor  201  is discharged with a constant current. When the capacitor  201  is discharged with the constant current, the voltage V 1  at the terminals of the capacitor  201  drops as shown in  FIG. 8 . The drop is equivalent to a value detected during the on period of the MOSFET  1 . 
     Thereafter, when the MOSFET  1  is turned off, the voltage VP 2  becomes positive. After a certain dead time elapses, the MOSFET  2  is turned on, and the capacitor  201  is charged with a constant current during the on period T 2 . When the voltage at the terminals of the capacitor  201  exceeds a base-emitter saturation voltage VBE 1  of the NPN transistor  204 , the NPN transistor  204  is turned on. Consequently, the PNP transistor  103  shown in  FIG. 1  is turned on, and the MOSFET  2  is turned off. 
     At this time, in case the currents flowing through the constant current circuits  400   a  and  400   b  shown in  FIG. 3  are set to an equal value, the absolute values of the change rates of the voltage at the terminals of the capacitor  201  observed during periods T 1  and T 2  respectively become equal to each other, and the widths of the changes in the voltage observed during the periods T 1  and T 2  respectively shown in  FIG. 8  become equalized to each other. In other words, a drop in the voltage at the terminals of the capacitor  201 , which drop is equivalent to a value detected during the on period T 1  of the MOSFET  1 , reaches the voltage VBE 1  during the same time as the on period T 1 . Consequently, the on time T 2  of the MOSFET  2  becomes nearly equal or substantially equal to the on time (T 1 ) of the MOSFET  1 . Eventually, a total loss caused by the MOSFETs and rectification diodes is minimized. 
     Since the currents flowing through the constant current circuits  400   a  and  400   b  shown in  FIG. 3  are set to the same value, only one constant current circuit  400   c  can be used instead to form a charge/discharge circuit  200   b , such as shown in  FIG. 4 . Unlike the charge/discharge circuit  200   a , the charge/discharge circuit  200   b  uses four diodes  403 - 406 . However, since the constant current circuit  400   c  alone is needed, the circuitry is simplified and the cost reduction is easily achieved. Similarly to the charge/discharge circuit  200   a  shown in  FIG. 3 , any of the constant current circuitries shown in  FIG. 5  to  FIG. 7  can be used as the constant current circuit  400   c.    
       FIG. 9  is a circuit diagram showing another embodiment  100   b  of the control unit  100 . In this embodiment, a charge/discharge circuit  300 , a capacitor  301 , a diode  302 , and an NPN transistor  303  are added to the control unit  100   a  shown in  FIG. 2 . In the circuitry shown in  FIG. 15 , the lighter the load connected to the DC output terminal of a DC-DC converter is, the shorter the on time T 1  of the MOSFET  1  becomes. Accordingly, the on time T 2  of the MOSFET  2  gets shorter. Eventually, a driving frequency gets higher, and the efficiency in conversion of the DC-DC converter becomes degraded. Therefore, the control unit  100   b  is designed to restrict the minimum value of the on time T 2  of the MOSFET  2  to a value Tmin. Thus, the driving frequency is restricted to a value 1/Tmin, and the degradation in the conversion efficiency of the DC-DC converter can be suppressed. 
       FIGS. 10 and 11  illustrate actions to be performed in the control unit shown in  FIG. 9 .  FIG. 10  shows waveforms signifying actions to be performed in a case where the on time T 1  of the MOSFET  1  is longer than the time Tmin.  FIG. 11  shows waveforms signifying actions to be performed in a case where the on time T 1  of the MOSFET  1  is shorter than the time Tmin. Incidentally, the waveforms shown in  FIG. 10  and  FIG. 11  represent the same voltages as those shown in  FIG. 8 . The voltage V 2  represents the voltage at the terminals of the capacitor  301 . In  FIG. 10 , the voltage VP 2  remains negative during the period T 1  during which the MOSFET  1  is on. The charge/discharge circuit  300  discharges the capacitor  301  until the voltage across the capacitor  301  drops to the forward voltage (−VF) across the diode  302 . 
     After the MOSFET  1  is turned off, when the voltage VP 2  is changed from a negative value to a positive value, the capacitor  301  is charged by the charge/discharge circuit  300  until the voltage at the capacitor becomes equal to the base-emitter saturation voltage VBE 2  of the transistor  303 . After the voltage at the capacitor  301  reaches the designated Tmin value, the transistor  303  is turned on. The timing when the transistor  204  is turned on next is, similarly to the one in the circuitry shown in  FIG. 2 , the timing when the voltage at the capacitor  201  reaches the base-emitter saturation voltage VBE 1 . Consequently, the on time T 2  of the MOSFET  2  is nearly equal to the on time T 1 . 
     On the other hand, as shown in  FIG. 11 , when the on time T 1  of the MOSFET  1  is shorter than the time Tmin, after the set time Tmin has elapsed since the MOSFET  1  is turned off, the transistor  303  is turned on. A charge time required for the voltage V 1  to be developed across the capacitor  201  is longer than a discharge time required therefor. The voltage V 1  is therefore higher than the base-emitter saturation voltage VBE 1  of the transistor  204 . Consequently, when the transistor  303  is turned on, the transistor  204  is turned on at the same time. The on time T 2  of the MOSFET  2  equals to the time Tmin, and the driving frequency is restricted to the frequency 1/Tmin. 
     As mentioned above, when the on time T 2  of the MOSFET  2  is restricted to the time Tmin, the on time T 1  of the MOSFET  1  is, as denoted by T 1 ′ in  FIG. 11 , shorter than the time Tmin. The current flowing through the MOSFETs  1  and  2  and the current flowing through the rectification diodes  7  and  8  become unbalanced. However, since the load is light, the increase in the loss becomes negligible. 
     Moreover, when the voltage VP 2  at the tertiary winding of the transformer  6  is positive, the charge/discharge circuit  300  charges the capacitor  301 . The time elapsing until the voltage at the terminals of the capacitor  301  changes from the forward voltage (−VF) across the diode  302  to the base-emitter saturation voltage VBE 2  of the NPN transistor  303  is equal to the time Tmin. In other words, after the time Tmin elapses, both the NPN transistors  303  and  204  are turned on. This causes the MOSFET  2  to be turned off. Moreover, when the voltage VP 2  is negative, the capacitor  301  is discharged by the charge/discharge circuit  300  until the voltage across the capacitor  301  becomes equal to the forward voltage (−VF) across the diode  302 . 
     The circuitry shown in  FIG. 3  can be used as the charge/discharge circuit  300 . Since the on time of the MOSFET  1  is shorter than the time Tmin when a light load is imposed, current flowing through the constant current circuit  400   b  is set to a larger value than that flowing through the constant current circuit  400   a . However, since the discharge time for the capacitor  301  need not be precise, a resistor or the like can substitute the constant current circuit. When the driving frequency to be restricted need not be precise, the charge/discharge circuit  300  can have the circuitry  300   a  shown in  FIG. 12  or the circuitry  300   b  shown in  FIG. 13 . 
     The present device can control the on times of the two MOSFETs so that the on times are nearly equal to each other. Consequently, the losses caused by the MOSFETs and rectification diodes can be reduced. Eventually, heat sinks that cool the MOSFET and rectification diodes can be downsized, and the switching power supply device can be made compact and economically. 
     While the present invention has been particularly shown and described with reference to preferred embodiment thereof, it will be understood by those skilled in the art that the foregoing and other changes in form and details can be made therein without departing from the spirit and scope of the present invention. All modifications and equivalents attainable by one versed in the art from the present disclosure within the scope and spirit of the present invention are to be included as further embodiments of the present invention. The scope of the present invention accordingly is to be defined as set forth in the appended claims. 
     This application is based on and claims priority to Japanese Patent Application 2006-305197 filed on Nov. 10, 2006. The disclosure of the priority application in its entirety, including the drawings, claims, and the specification thereof, is incorporated herein by reference.