Abstract:
A synthesizer for generating a desired chirp signal has M parallel channels, where M is an integer greater than 1, each channel including a chirp waveform synthesizer generating at an output a portion of a digital representation of the desired chirp signal; and a multiplexer for multiplexing the M outputs to create a digital representation of the desired chirp signal. Preferably, each channel receives input information that is a function of information representing the desired chirp signal.

Description:
The United States Government has rights in this invention pursuant to Department of Energy Contract No. DE-AC04-94AL85000 with Sandia Corporation. 
    
    
     CROSS REFERENCE TO RELATED APPLICATIONS 
     (Not Applicable) 
     BACKGROUND OF THE INVENTION 
     Many radar applications, including synthetic aperture radar (SAR), require fine range resolution. A simplistic method of achieving fine range resolution is to transmit a very narrow pulse and adequately sample the returned echo. This method, while simple in theory, has some practical limitations: very high peak transmitter power and very high bandwidth for an analog-to-digital (A/D) converter. 
     Pulse compression is one known radar technique for reducing peak transmitter power while maintaining a fixed average transmitter power by coding the transmitted waveform in such a manner as to be able to decode the received echo into the various constituents of the transmitted code. 
     Some pulse compression schemes, such as binary and polyphase codes, literally change the phase of the transmitted signal at regular sub-pulse intervals. The length of these sub-pulse intervals determines the achievable range resolution. 
     Another pulse compression scheme involves continuously varying the phase of the transmitted waveform. When this phase variation is quadratic (and, therefore, the frequency variation is linear), the method is called linear FM pulse, or chirp pulse compression. Such a waveform is described by: 
     
       
           y ( t )=sin({2π( f   0   t +0.5 kt   2 +φ 0 )}, 0 ≦t ≦τ  eq. 1 
       
     
     where k is the linear chirp rate, τ is the pulse length, and f 0  and φ 0  are the starting frequency and phase of the waveform, respectively. 
     A preferred technique for implementing chirp pulse compression utilizes the direct-digital-synthesis (DDS) circuitry of prior art FIG. 1 to generate y(f). Conventional frequency accumulator  2  includes an adder  32  in series with a register  34  and a feedback loop  36  from the output of register  34  to the input through adder  32 . This circuit accumulates, or integrates, the chirp rate constant, k, and adds it to a programmed starting frequency f 0  (which had been loaded into register  34 ) to provide an output f(t)=f 0 +kt, the instantaneous frequency. This output is used as the input to phase accumulator  4  where the linear frequency term is integrated to provide an output φ(t)=f 0 t−kt/2+kt 2 /2+φ 0 , the instantaneous phase. The combination of these first two accumulators is collectively known as the phase generator PG. The output of phase accumulator  4  is applied as an address to a mapping device such as look-up table ROM  6  which contains one cycle of a sine. The resulting output of ROM  6  takes the form of equation 1. This circuit is recognizable as the digital portion of a conventional DDS (formed by phase accumulator  4 , look-up table ROM  6 ), with the addition of a frequency accumulator  2  at the input to provide for the changing frequency of the chirp generator. The output of look-up table ROM  6  is fed through a D/A converter  7  to provide an analog signal for transmission by a radar. The phase error correction look up table (PEC LUT)  8  is an important part of most modem chirp synthesizers that compensates for non-linearity&#39;s in RF components of a radar in a manner known to those of ordinary skill in the art. 
     The advantages of this design include a waveform length independent of hardware configuration, ease of changing waveform parameters, a capability of generating a continuous, constant frequency (CW) sinusoid, and a small part count. To change waveform parameters, all that is required is to change the values for the starting frequency f 0 , the starting phase φ 0 , and the chirp rate k, all of which values are stored in registers. Pulse length, τ, is also programmable. If desired, these parameters may be changed on a pulse by pulse basis. Since a chirp radar system requires two waveforms, one for use during transmit and a second for use during receive, the wave form synthesizer (WFM) chirp generator of FIG. 1 may be used with the starting frequency and pulse duration for each pulse loaded into different registers to permit independent specification of the transmit and receive waveforms. 
     The implementation of the chirp generator of FIG. 1 may take many embodiments. However, this generator, and DDS systems in general, are limited in output to about 50% of the clock frequency because of the Nyquist sampling theorem. To generate an output frequency on the order at a sampling rate of 1 GHz, prior art systems have utilized expensive, high power, custom made GaAs integrated circuits. (See, for example, B. Redmund et al.,  A  500  MHz Phase Generator for Synthetic Aperture Radar Waveform Synthesizers , Sandia National Laboratories publication SAND 91-0966C, July 1991.) 
     SUMMARY OF THE INVENTION 
     It is an object of this invention to provide a high frequency chirp circuit utilizing commercially available components. This and other objects and advantages of the invention may be realized and attained as particularly pointed out in the appended claims. 
     To achieve the foregoing and other objects, and in accordance with the purpose of the present invention, as embodied and broadly described herein, the present invention may comprise a synthesizer for generating a desired chirp signal comprising M parallel channels, where M is an integer greater than 1, each channel including a chirp waveform synthesizer generating at an output a portion of a digital representation of the desired chirp signal; and a multiplexer for multiplexing the M outputs to create a digital representation of the desired chirp signal. Preferably, each channel receives input information that is a function of information representing the desired chirp signal. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The accompanying drawings, which are incorporated in and form part of the specification, illustrate an embodiment of the present invention and, together with the description, serve to explain the principles of the invention. 
     FIG. 1 shows a prior art chirp synthesizer. 
     FIG. 2 shows the Multiplexed Waveform Synthesizer(MWFS) 
     FIG. 3 shows the initializer for the MWFS. 
     FIG. 4 shows an embodiment of the multiplexer of the invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     The theory behind this invention is relatively simple. In the prior art single channel chirp synthesizer of FIG. 1 with registers P (phase), F (frequency), and K (chirp rate), the register values as a function of clock cycle (n) are given by: 
     
       
           K   s =ChirpRate  eq. 2 
       
     
     
       
           F   s ( n )= F   s ( 0 )+ K   s   n   eq. 3 
       
     
       P   s ( n )= P   s ( 0 )+ F   s ( 0 ) n +½ K   s ( n   2   −n )  n =0,1,2,3, . . .   eq. 4 
     
       
           y ( n )=sin[ P ( n )]  eq. 5 
       
     
     where F s ( 0 ) represents the desired start frequency of the chirp waveform and y(n) gives the output of the sine ROM  6 . As is well known in the art, D/A converter  7  converts the digital data stream P(n) to P(t), a continuous time function where t=n*T s , and T s  is the sample period and S=1/T s  is the sample frequency of the D/A converter. 
     The objective of the invention is to combine M parallel channels of chirp waveform synthesis, with each channel running at only S/M, in such a way that the input values for starting phase, frequency and chirp rate are identical to those applied to the WFS of FIG. 1, and the output of the combined channels is identical to the output of the prior art MWS of FIG. 1 . 
     This objective is achieved by the multiplexed waveform synthesizer (MWFS) of FIG.  2 . In this scheme the M-to-1 multiplexer  20  receives data from all M channels every M clock cycles and outputs this data sequentially as y(n) through a D/A converter  24  to form the analog radar signal. Each channel m of the MWFS is similar to the WFS of FIG. 1 as they include a frequency accumulator  2 (m), a phase accumulator  4 (m), and a sine look up table  6 (m). However, each channel processes its part of the output at a rate of S/M, which enables each channel to be implemented in low cost CMOS logic. Only the output stages of multiplexer  20  and D/A  24  operate at the much higher rate S. 
     In accordance with the invention, if there are M parallel channels of chirp waveform generators, then the output of the accumulators of the m th  channel will be denoted: 
     
       
           F   m ( n )= F   m ( 0 )+ K   m   n m =0,1,2 , . . . M −1  eq. 6 
       
     
     
       
         
           
             
               
                 
                   
                     
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     where F m ( 0 ) and P m ( 0 ) are the starting values for the frequency and phase accumulators, respectively, of the m th  channel. 
     In order for the MWFS to produce the same results as the single channel synthesizer, we must have: P m (n)=P s (M·n+m) where P s (n) is the phase output of the prior art single channel WFS of FIG. 1, P m (n) is the output of the m th  channel of the MWFS, and M is the number of channels in the MWFS. A little algebra reveals the necessary terms.            P   m          (   n   )       =         P   s          (   0   )       +         F   s          (   0   )            (     Mn   +   m     )       +       1   2                       K   s          [         (     Mn   +   m     )     2     -     (     Mn   +   m     )       ]                                  
     or                       P   m          (   n   )       =                    P   s          (   0   )       +     m                     F   s          (   0   )         +       1   2                       K   s          (       m   2     +   m     )         +                                n        [       M        (         F   s          (   0   )       +       K   s        m       )       +       1   2          M   2          K   s       -       1   2          MK   s         ]       +       1   2          K   s            M   2          (       n   2     -   n     )                                          
     which is in the same form as eq. 7. Accordingly, the starting values of the accumulators of the M channels are:                  P   m          (   0   )       =                    P   s          (   0   )       +     m                     F   s          (   0   )         +       1   2                       K   s          (       m   2     -   m     )                   eq   .              8                   F   m          (   0   )       =       M        (         F   s          (   0   )       +       K   s        m       )       +       1   2          M   2          K   s       -       1   2          MK   s                 eq   .              9                                K   m   =K   s   M   2   eq. 10. 
     A major aspect of this invention is how the frequency and phase accumulators of each channel are efficiently loaded with the appropriate starting values. This task is the responsibility of the initializer circuit  50  of FIG.  3 . 
     Initializer  50  has sections for generating phase, frequency, and chirp rate information, and a finite state machine (FSM)  60  for providing clock enable signals and controlling the system. Registers  53 - 57  control the flow of data through initializer  60  as their clock enable ports are activated by FSM  60  in a conventional manner to those of ordinary skill in the art. In a similar manner, the select lines (not shown) of the multiplexers are also controlled by FSM  60  A multiplexer  52  receives this data and passes it through register  54  to point A on FIG.  2 . 
     The starting phase information, P s ( 0 ), is applied directly to multiplexer  52  through register  53 . Referring to eqs. 4 and 8 above, the starting value P m ( 0 ) for the phase register  2 (m) of the m th  channel is exactly the value of the single channel WFS after it is clocked m times. Therefore, the phase register starting values for the each channel are set by loading the single channel starting values F s ( 0 ) and P s ( 0 ) into the frequency and phase accumulators of all the channels under the control of the FSM and then clock enabling each channel m that number of times (i.e., channel  2  is clocked twice; channel  12  is clocked 12 times). 
     The starting frequency information, F s ( 0 ), is applied to multiplexer  52  through a combination of multiplexers and gates, as illustrated. The first term of eq. 9 can be generated by loading all the frequency accumulators with M*F s ( 0 ) and then adding M*K s , m times. The multiplication by M is performed in the initializer&#39;s arithmetic logic; the addition is done in the frequency accumulator of the channel. The next two terms of eq. 9 are similarly produced in the arithmetic logic of the initializer and accumulated into the frequency accumulators  2 (m) of the channels. 
     The final step to initialize the channels is to form the correct value for K m =M 2 * K s . This value is computed in the arithmetic logic of the feedback loop  58  of the initializer and placed on the K bus of the MWFS. 
     Chirp synthesis begins when all accumulators of the MWFS are turned on. A final note should be made about the accumulator for the PEC LUT. It is loaded initially with F s ( 0 ) from FSM  60 , and it has a constant value of M*K s  on the accumulator input during chirp synthesis that is provided by output B of initializer  60   
     There is a significant advantage to having the number of channels M=2 k , where k is a positive integer (M=2,4,8,16, . . . ). In this case, the multiplications referred to above may be accomplished by merely shifting the bits in a register in a manner well known in the art. For a different number of channels, more complicated circuitry must be utilized to perform the multiplications 
     To prove the concept of this invention, a computer program was written to show that the outputs of the m channels would combine to form the same output as a single high speed channel. Even though no single channel had a signal that was recognizable as the desired output, the combination of all channels was shown to be identical to the output from a single high speed channel. 
     A preferred embodiment of a multiplexed waveform synthesizer (MWFS) in accordance with this invention has been constructed on a single VME card that uses direct digital synthesis (DDS) techniques for chirp generation. (A VME card is a card utilized with a VersaModule Eurocard bus, a 32-bit bus developed by Motorola, Signetics, Mostek and Thompson CSF that is widely used in industrial, commercial and military applications.) The MWFS is designed to support pulse compression in Synthetic Aperture Radar (SAR) by generating chirps at 1000 megasamples/second to provide 50 MHz to 450 MHz sinusoidal outputs. 
     The preferred embodiment MWFS contains M=2 4 =16 WFS channels implemented in two identical CMOS Field Programmable Gate Arrays (FPGA&#39;s) (not shown). The programming of FPGAs to implement digital circuitry is a routine manner for those of ordinary skill in the art and does not form part of this invention. This circuit could also be implemented as a physical circuit using standard manufacturing techniques, but the FPGA embodiment does offer the additional benefit of flexibility, so that different configurations may be utilized for different missions. 
     As shown in FIG. 4, eight channels (M=2 3 ) are implemented in each of two FPGAs. Initializer  60  of FIG. 3 is also implemented in at least one FPGA. Each WFS channel runs at S/M=1000 MHz/16=62.5 MHz. FIG. 2 shows that each WFS channel also includes a four quadrant sine ROM  6 (m) with 12 phase input bits and 9 magnitude output bits. In this embodiment, the individual WFS channels are arranged in pairs and each pair is multiplexed together through 2-1 mux  40  (m/ 2 ) to form a 125 MHz data stream before leaving the FPGA. These 125 MHz streams are then multiplexed together in 8-1 mux  20  which is implemented using commercial ECL logic chips as shown in FIG. 4 to form the 1000 MHz data stream. The output of eight-to-one multiplexer  20  feeds a Digital to Analog Converter  24  to generate the desired analog chirp signal. 
     As implemented, each WFS channel includes 36 bit frequency and phase accumulators that run at 62.5 MHz. Each of these accumulators is preferably implemented as a 6 way pipeline in order to achieve the desired 62.5 MHz clock rate. In order to fit 8 sine ROM&#39;s  6 (m) into each FPGA, it was necessary to use a standard Sine Table compression scheme similar to the Multiplying Sine ROM. Implementation of a look up table for this invention is a routine matter of design to those of ordinary skill in the art. 
     As mentioned earlier, the WFS channels are split between two FPGA&#39;s and both these FPGA&#39;s must receive the same information from the VME bus. To simplify programming and save access time the FPGA&#39;s have been mapped so that the same register in both FPGA&#39;s can be accessed with a single VME cycle. All of the waveform synthesizers preferably would have been implemented on a single device if a larger FPGA had been available. 
     The ECL 8-to-1 multiplexer  20  required phase locked loops (PLL&#39;s) and precisely controlled trace lengths to function properly. A diagram of multiplexers  40  and  20  is shown in FIG. 4 to include four levels of 2-stage of multiplexing (mu 1 , mu 2 , mu 3 , mu 4 ), with the inputs for each stage being selected to maximize the time between signal inputs for each stage. For example, channels  3  and  11  are combined in multiplexer  3  of stage mu 1  while channels  7  and  15  are combined in multiplexer  7 . The outputs of these two multiplexers are combined in multiplexer  81  of stage mu 2 . As shown in FIG. 4, all odd numbered channels are combined in multiplexer  91  of stage mu 3 , and all even numbered channels are combined in multiplexer  92 . Only output multiplexer  99  of stage mu 4  and D/A converter  24  are clocked at the 1 GHz sample rate S. 
     The input values of start phase, frequency, and chirp rate for this embodiment of the invention have been loaded into registers  62 ,  64 , and  66  in the same manner as they would have been for a single channel synthesizer. The output chirp waveform from D/A converter  24  is identical to the output of a single channel synthesizer. However, this embodiment generated the signal at a 1 GHz sample rate utilizing mostly CMOS circuitry instead of the expensive GaAs circuitry that would be required for a single channel circuit. 
     The particular sizes and equipment discussed above are cited merely to illustrate a particular embodiment of this invention. It is contemplated that the use of the invention may involve components having different sizes and shapes as long as the principle, multiplexing low-speed waveform synthesizers to create a high speed chirp, is followed. For example, if the accumulator widths or sine ROM sizes were to change, the claims of this document would still apply. It is intended that the scope of the invention be defined by the claims appended hereto.