Abstract:
The invention discloses a novel equalization system for reducing the deleterious effects of crosstalk on signals received at a modem, with particular regard to QAM signals transmitted over copper twisted pairs. The equalization system employs the common combination of a linear equalizer followed by a decision-feedback equalizer (DFE). However, instead of aiming for perfect equalization of channel distortion, the overall frequency response of the channel plus the linear equalizer is skewed such that higher frequencies are attenuated relative to lower ones. More generally, the spectral regions where crosstalk is strongest are attenuated, which reduces the crosstalk noise present at the input to the DFE at the expense of introducing inter-symbol interference (ISI). Fortunately, most DFEs are capable of handling the added ISI, leading to data decisions that are relatively unaffected by crosstalk noise.

Description:
FIELD OF THE INVENTION 
     The present invention is concerned with the reception of electronic signals and particularly relates to a method and apparatus for reducing the effects of crosstalk on received subscriber loop signals. 
     BACKGROUND OF THE INVENTION 
     A signal travelling along an unshielded copper twisted pair undergoes frequency-dependent attenuation due to the transmission characteristics of the cable. This distortion can be modelled as a slowly time-varying filtering operation applied to the transmitted signal. The relative variations in the attenuation at different frequencies cause phase shifts in the transmitted waveform and, in digital transmission systems, ultimately lead to spreading of the transmitted symbols into adjacent symbol intervals, a phenomenon known as inter-symbol interference (ISI). This results in the receiver committing errors when data decisions are made based on the distorted received signal. 
     In order to compensate for the distortive effects of cable loss, it is customary to employ a linear equalizer, possibly even an adaptive one, followed by a decision-feedback equalizer (DFE). The linear equalizer usually consists of a filter, either digital or analog, which has a frequency response equal to (or close to) the inverse response of the twisted pair “channel”. Therefore, if the transmitted signal is attenuated by a downwards sloping frequency response as it travels along the twisted pair, then the frequency response of the linear equalizer will have an upwards sloping shape. 
     The purpose of linear equalization is to equalize, or “flatten”, the overall channel response affecting the transmitted signal as it arrives at the decision-feedback equalizer. This linear equalization process has the effect of removing much, but not all, of the ISI corrupting the transmitted signal. The DFE is then used for removing any remaining ISI and for making data decisions on the received digital signal. Much theory has been developed around the problems of channel equalization and decision-feedback equalization, and various implementations have proven successful in the case of a digital signal transmitted along isolated media. 
     However, when a telephone company considers delivering digital signals to a plurality of subscribers over a copper twisted pair infrastructure, it is often the case that multiple twisted pairs are bundled together for at least part of the journey between a central office and the subscribers. As a result of poor shielding provided by the thin layer of insulant surrounding each copper wire, electromagnetic fields may be induced by one wire into other wires in the bundle, creating an effect known as crosstalk. 
     In general, crosstalk couples more at higher frequencies and therefore after a long voyage along a twisted pair, the effect of crosstalk on a transmitted signal will be most noticeable at high frequencies. A serious problem then occurs if the linear equalizer in the receiver boosts the high frequencies in an attempt to equalize the channel: the effects of crosstalk noise, which is stronger at higher frequencies, are actually enhanced. 
     At the DFE, therefore, the received signal equalized in accordance with prior art techniques has reduced ISI but possibly increased crosstalk noise. When crosstalk noise becomes the dominant cause of distortion, as when many twisted pairs are bundled together, the decision-feedback equalizer is incapable of making correct data decisions, with obvious deleterious consequences. 
     SUMMARY OF THE INVENTION 
     It is an object of the present invention to mitigate or obviate one or more disadvantages of the prior art. 
     Therefore, the invention may be summarized according to a first broad aspect as an equalization system for equalizing an input signal corrupted by inter-symbol interference and crosstalk noise, the equalization system comprising: a linear equalizer for filtering the input signal and producing an intermediate signal having reduced crosstalk noise and reduced inter-symbol interference; and a decision-feedback equalizer connected to the linear equalizer, for filtering the intermediate signal and further reducing the inter-symbol interference contained therein. 
     According to a second broad aspect, the invention may be summarized as an equalization system for producing an output data signal from an input signal corrupted by channel distortion and crosstalk noise, the crosstalk noise having a frequency characteristic, the channel distortion causing inter-symbol interference, the equalization system comprising: an adaptive linear equalizer for filtering the input signal and producing an intermediate signal having reduced crosstalk noise and reduced inter-symbol interference, wherein the adaptive linear equalizer has a frequency response controllable by a first error signal; a decision-feedback equalizer connected to the adaptive linear equalizer, for filtering the intermediate signal and further reducing the inter-symbol interference contained therein, thereby to produce the output data signal; an offset filter connected to the adaptive linear equalizer, for filtering the intermediate signal in accordance with a selectable offset frequency response and producing an offset-filtered signal, wherein the offset frequency response is selected to resemble the crosstalk frequency characteristic; and an error calculation block connected to the decision-feedback equalizer, to the offset filter and to the adaptive linear equalizer, for measuring a characteristic of the difference between the offset-filtered signal and a delayed version of the output data signal, thereby to produce the first error signal. 
     The invention may be summarized according to a third broad aspect as a method of equalizing an input signal corrupted by inter-symbol interference and crosstalk noise, comprising the steps of: filtering the input signal with a linear equalizer, thereby to produce an intermediate signal having reduced crosstalk noise and reduced inter-symbol interference; and filtering the intermediate signal with a decision-feedback equalizer, thereby to further reduce the inter-symbol interference in the intermediate signal. 
     In a method of equalizing an input signal corrupted by inter-symbol interference caused by cable loss varying in the frequency domain, said method consisting of filtering the input signal with a linear equalizer having a frequency response which compensates for the inter-symbol interference, the invention may be summarized according to another broad aspect as the improvement wherein the frequency response of the linear equalizer is deliberately made different from the frequency response which fully compensates for the inter-symbol interference so as to compensate for crosstalk, the remaining inter-symbol interference being further reduced by filtering with a decision-feedback equalizer. 
     According to yet another broad aspect, the invention may be summarized as a receiver used for producing a plurality of streams of digital data from a received analog input signal containing an analog message signal carrying the digital data, the message signal being centered about a carrier frequency and corrupted by channel distortion and crosstalk noise having respective frequency characteristics, the channel distortion causing inter-symbol interference, the receiver comprising: a bandpass filter for receiving the analog input signal and extracting the analog message signal therefrom; an analog-to-digital converter connected to the bandpass filter for converting the analog message signal to an intermediate digital signal; a demodulator connected to the analog-to-digital converter for creating a plurality of digital output signals from the intermediate digital signal; at least one decision-feedback equalizer connected to the demodulator for filtering the digital output signals and creating the digital data streams therefrom; and adaptive equalization means placed at one or more points between the bandpass filter and the decision-feedback equalizer, the equalization means having a selectable overall frequency response; wherein the overall frequency response of the equalization means is selected to resemble the difference between the inverse of the frequency characteristic of the channel distortion and the frequency characteristic of the crosstalk noise. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The preferred embodiment of the present invention will now be described with reference to the following figures, in which: 
     FIG. 1 is a high-level block diagram of a transmission system and particularly a receiver in accordance with the preferred embodiment of the present invention; 
     FIG. 2A is a high-level block diagram of a receiver in accordance with a first alternate embodiment of the present invention; 
     FIG. 2B is a detailed block diagram of part of the receiver in FIG. 2A; and 
     FIG. 3 is a detailed block diagram of part of a receiver in accordance with a second alternate embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     With reference to FIG. 1, there is shown a transmitter  50  for transmitting an analog version of a digital signal across a channel  60  to an inventive receiver  100 . The receiver  100  would typically form part of a modem at customer premises equipment or at a network unit interfacing with twisted pairs leading to individual subscribers. 
     At the receiver  100 , a received analog signal  151  enters an analog bandpass filter  101 , an analog equalizer  102 , a variable gain amplifier (VGA,  103 ) and an analog-to-digital converter (ADC,  104 ) prior to entering a demodulator  105 . The VGA  103  is controlled by a control signal  150  from the demodulator  105  and produces a level-controlled signal  154  leading to the ADC  104 . 
     The demodulator  105  is preferably a quadrature demodulator and produces two baseband digital signals, namely, an in-phase signal  156 A and an in-quadrature signal  156 B, which are fed to respective identical (linear) digital equalizers  106 A,B. The demodulator  105  contains an automatic gain control (AGC) function which controls the VGA  103  via control signal  150  that varies in accordance with the difference between an estimate of the combined power of the baseband digital signals  156 A,B and a desired value. 
     The two baseband digital signals  156 A,B follow parallel paths through respective digital equalizers  106 A,B and decision-feedback equalizers (DFEs)  107 A,B. In the embodiment of FIG. 1, the digital equalizers  106 A,B are fixed digital filters and could be finite impulse response (FIR) filters (as is preferred) or infinite impulse response (IIR) filters, producing respective equalized digital signals  157 A,B. The DFEs  107 A,B strive to eliminate inter-symbol interference present in the equalized digital signals  157 A,B exiting the digital equalizers  106 A,B, producing respective data streams  158 A,B that are used by additional customer premises or telco components connected to the receiver, such as buffers, computer memories and the like. 
     While the receiver  100  in FIG. 1 appears to resemble a prior art receiver, there has been a significant modification to the composite frequency response provided by the analog equalizer  102  and digital equalizers  106 A,B. Assuming that the channel is afflicted with a frequency response  2  having a particular slope, a prior art receiver would apply a composite analog and digital frequency response  4  having a slope that is inversely proportional to the slope of the channel frequency response  2 . 
     In the preferred embodiment of the present invention, however, the digital equalizers  106 A,B (at baseband) and the analog equalizer  102  (at passband) apply a composite frequency response  6  which is similar to the prior art frequency response  4  but is milder in slope, i.e., the inventive receiver has a composite frequency response which deviates from the “ideal” channel equalization response by providing higher attenuation at higher frequencies. The offset frequency response, defined as the difference between the “ideal” response  4  and the inventive response  6 , desirably rises by 1 to 20 decibels across the spectral region of interest, which ranges between 1 MHz and 30 MHz in the case of QAM transmission over copper twisted pairs. 
     In operation, the received analog signal  151  is an analog version of a digitally modulated signal, preferably a quadrature amplitude modulation (QAM) signal of order  4 ,  16 ,  64  or  256 , having side lobes centered about a carrier frequency in the range of 1 MHz to 30 MHz. While these limits represent the most practical operational range for local loop transmission systems, it is within the scope of the present invention to use any order of QAM or in fact any other digital modulation scheme in any frequency range. For example, the present invention is also applicable to systems using carrierless amplitude and phase (CAP) modulation in the 1-30 MHz frequency range. 
     The analog bandpass filter  101  selects only those portions of the frequency spectrum of the received analog signal  151  which are of interest, i.e., the spectral region surrounding the carrier frequency. The analog equalizer  102  then performs another filtering operation on signal  152  exiting the analog bandpass filter  101 . The frequency response of the analog equalizer  102  is matched to partially compensate for losses due to the twisted pair cable that can be easily estimated or are known in advance. 
     The gain of the level-controlled signal  154  output by the VGA  103  is adjusted according to the power (or energy or magnitude) level of the baseband digital signals  156 A,B as demodulated by the quadrature demodulator  105 . This ensures that a relatively constant signal gain is maintained at the input to the ADC  104 . The quadrature demodulator  105  is a component commonly used in the art for producing the in-phase and in-quadrature baseband demodulated signals  156 A,B from the (quadrature modulated) level-controlled signal  154 . 
     The operations performed by digital equalizer  106 A and DFE  107 A are identical to those performed by digital equalizer  106 B and DFE  107 B, and thus it is sufficient to continue describing the invention with reference only to the “B” phase involving digital equalizer  106 B and DFE  107 B. 
     Digital equalizer  106 B partially compensates for cable loss and has a frequency response (at baseband in the digital domain) which, in combination with the frequency response (at passband in the analog domain) of the analog equalizer  102 , approximates the inverse channel response of the copper twisted pair offset by a mild decay, as represented by frequency response  6 . This effectively attenuates higher frequencies relative to lower ones, and therefore reduces the effect of crosstalk noise, which is known to be predominant at higher frequencies. 
     However, applying a frequency response such as frequency response  6  instead of the “ideal” frequency response  4  causes imperfect channel equalization and introduces additional inter-symbol interference into the equalized digital signal  157 B fed to the DFE  107 B. The DFE  107 B then strives to remove the inter-symbol interference due both to cable distortion not compensated for by the equalizers  102 ,  106 B and to distortion caused by the compensation gradient (offset slope) of the composite frequency response  6  of these two filters. 
     The maximum acceptable offset slope steepness (or minimum acceptable slope steepness of the frequency response  6 ) is reached when the total inter-symbol interference becomes irreparable even by a DFE. In the case of QAM, where the side lobes centered about the carrier frequency are symmetrical, excessive attenuation at higher frequencies can be partially compensated for by emphasizing the lower band. (This compensatory biasing effect occurs naturally in the DFE algorithm as it strives to make correct decisions based on the equalized digital signal  157 B.) 
     It is to be understood that the offset introduced into the composite frequency response  6  of the analog equalizer  102  and the digital equalizers  106 A,B need not resemble a linearly sloping characteristic. Instead, the offset, i.e., the difference between the ideal and inventive composite frequency responses, may have any characteristic that reduces the effect of crosstalk noise, which is especially dominant at higher frequencies, with the additional constraint that the resultant inter-symbol interference still be cancellable by the DFE. Nevertheless, it is preferable to employ some relatively simple shape for the offset. 
     While the preferred embodiment of the present invention has been described and illustrated, it will be apparent to one skilled in the art that numerous modifications and variations are possible. For example, analog-to-digital conversion may be executed at points other than at the input to the demodulator  105 , such as prior to the VGA  103  or subsequent to demodulation. Moreover, passband filtering performed by the analog equalizer  102  may be omitted or, alternatively, baseband filtering executed by the digital equalizers  106 A,B may be left out. 
     It is also possible to build on and improve the inventive receiver in FIG. 1 by adapting the equalizers  106 A,B so as to track time-varying cable characteristics while continuing to provide a compensatory frequency response which reduces crosstalk noise. To this end, FIG. 2A shows a receiver  200  seen to comprise all of the components of the receiver  100  in FIG. 1 in addition to offset filters  208 A,B and error calculation blocks  209 A,B. 
     It is noted that the “A” and “B” phases are structurally identical. In the case of phase “B”, now considered, the offset filter  208 B applies a desired offset shape  10  to the equalized digital signal  157 B, producing an offset filtered signal  261 B. It is preferable that the shape of the frequency response  10  of the offset filter  208 B be made to resemble, as much as possible, the spectral characteristic of the induced crosstalk noise, e.g., an upwards-sloping curve. Data stream  158 B is also fed to the error calculation block  209 B, which supplies a first error signal  263 B to the digital equalizer  106 B. A second error signal  231 B is fed to the digital equalizer  106 B directly from the DFE  107 B. 
     The digital equalizer  106 B then adapts its coefficients to minimize either the first error signal  263 B or the second error signal  231 B, resulting (upon convergence) in a composite frequency response  8  of the analog equalizer  102  and the digital equalizer  106 B which has a shape that is offset from the “ideal” channel equalization response by the user-defined offset frequency response  10 . A key advantage of this first alternate embodiment is that the amount of crosstalk noise can be controlled while allowing continuous adaptation of the composite frequency response to account for channel variations. 
     In order to gain a more complete understanding of this embodiment, it is useful to describe in further detail the structure of the receiver in FIG. 2A, and particularly that of box  200  surrounding the digital equalizer  106 B, the DFE  107 B, the offset filter  208 B and the error calculation block  209 B, as expanded in FIG.  2 B. 
     Structurally, the digital equalizer  106 B is shown as an adaptive FIR filter with a tapped delay line  210 , a series of multipliers  211  and respective coefficients  212 , an adder  213  and a control block  214 . The tapped delay line  210  consists of delay elements that retard the baseband digital signal  156 B by one sample period each. At the output of each delay element, the resultant signal is tapped and enters a respective multiplier  211 . While the tapped delay line  210  is shown as having only four taps in the interest of simplicity, it is most preferable to use anywhere from 24 to 32 taps, and a greater or smaller number may be used if desired. 
     At the control block  214 , the values of the first and second error signals  263 B and  231 B are passed to a control algorithm which adjusts the values of the coefficients  212  multiplying the outputs of the delay elements. (Generation of the first and second error signals  263 B and  231 B is discussed below.) The adder  213  adds the delayed and scaled signal values, thereby producing the equalized digital signal  157 B. 
     At the input end of the DFE  107 B, the equalized digital signal  157 B enters an adder  220  followed by a slicer  221 . The adder  220  adds a feedback signal  222  to the equalized digital signal  157 B producing a signal  229 . The slicer  221  makes a data decision based on the value of the signal  229  output by the adder  220 , resulting in data stream  158 B. In this way, the equalized digital signal  157 B is adjusted by the feedback signal  222  prior to the decision-making process. 
     The feedback signal  222  is an adaptively FIR-filtered version of the data stream  158 B and, accordingly, the DFE  107 B also comprises a tapped delay line  223  (accepting data stream  158 B), a plurality of multipliers  224  and corresponding coefficients  225 , an adder  226  and a control block  227  for adjusting the values of the multipliers  224 . Again, the number of taps in the tapped delay line  223  preferably lies between  24  and  32 , but can be greater or smaller if this leads to improved performance. 
     Still considering the DFE  107 B, the output  229  of the adder  220  passes through a delay block  217  and subsequently enters a comparator  216 . Also entering the comparator  216  is the output of the slicer  221 , namely data stream  158 B. The comparator  216  then measures a function (e.g., the absolute value or square) of the difference between the delayed samples and the decisions made by the slicer. The delay block  217  provides a time delay which compensates for the delay incurred by the slicer  221  in making a decision. 
     The output  231 B of the comparator  216  is then fed to the control block  227  in the DFE  107 B as well as to the control block  214  in the digital equalizer  106 B. The control block  227  in the DFE  107 B runs a conventional control algorithm to adjust the values of the coefficients  225  multiplying the outputs of the delay elements in the tapped delay line  223 . 
     Considering now the offset filter  208 B, it is shown in FIG. 2B as a simple FIR filter acting on the equalized digital signal  157 B received from the digital equalizer  106 B. The offset filter  208   b  could also be an IIR filter. In the style of an ordinary FIR filter, a plurality (preferably between 24 and 32) of coefficients  240  simultaneously multiply delayed versions of the equalized digital signal  157 B, the products being added together to form the output signal  261 B fed to the error calculation block  209 B. In the embodiment of FIG. 2B, the values of the coefficients  240  of the offset filter  208 B are kept fixed and provide an offset frequency response which resembles that of the crosstalk frequency characteristic, e.g., having an upward slope. 
     The output  261 B of the offset filter  208 B then enters a comparator  218  in the error calculation block  209 B. The comparator  218  also accepts the output  262  of another delay block  215 , which itself is fed by data stream  158 B. The comparator measures the absolute value (or power, etc.) of the difference between signals  261 B and  262  and feeds the resultant first error signal  263 B back to the control block  214  of the digital equalizer  106 B. For reasons to be discussed below, it is preferable that the delay applied by the delay block  215  be substantially identical to the delay introduced by the offset filter  208 B minus the delay of the DFE  107 B. 
     Operation of the inventive receiver depicted in FIGS. 2A and 2B is now considered, and can be broken down into three major steps. Again, in the interest of simplicity, only the “B” phase is considered, but it is to be understood that analogous treatment is applicable to the “A” phase. 
     (A) Startup 
     At startup, the coefficients  240  of the offset filter  208 B are given values which provide an upward slope or any other shape approximating the crosstalk behaviour on the particular transmission medium in question. In this first alternate embodiment, these coefficients will remain fixed throughout receiver operation. 
     (B) Initial Adaptation 
     During initial adaptation, control block  227  in the DFE  107 B adapts the DFE coefficients  225  based on the second error signal  231 B from the comparator  216 . (Internal adaptation of the DFE  107 B is achieved using standard algorithms and need not be described in further detail.) While waiting for the DFE  107 B to stabilize, control block  214  in the digital equalizer  106 D is programmed to adapt the coefficients  212  based only on the first error signal  263 B received from the error calculation block  209 B. In this way, any transient error values of the error signal  231 B output by the DFE  107 B will not affect adaptation of the digital equalizer  107 B. 
     When the coefficients  212  of the digital equalizer  106 B are close to convergence, it will be apparent that the power (or energy or absolute value, etc.) of the first error signal  263 B will be close to a minimum. It follows that the coefficients of the digital equalizer  106 B will provide a frequency response which, in combination with the frequency responses of the offset filter  208 B and analog equalizer  102 B, closely tracks the inverse channel response. At the same time, the DFE  107 B operates on samples which have reduced crosstalk, leading to better decisions, and hence a smaller value for the first error signal  263 B, ending up in convergence of the coefficients  211 . 
     The delay of delay block  215  is preferably equal to the delay of the offset filter  208 B minus the delay of the DFE  107 B. This permits the comparator  218  to deal with samples and with the data decisions corresponding to those samples. 
     (C) Steady-State Operation 
     Once the coefficients  212  of the digital equalizer  106 B have reached a point beyond which further convergence is impossible, e.g., when the power of the first error signal  263 B cannot be further reduced, then additional refinements can be obtained by the control block  214  switching to the second error signal  231 B as output by the comparator  216  in the DFE  107 B. 
     However, careful consideration of the block diagram in FIG. 2B reveals that if no attention is paid to the error signal  263 B provided by the error calculation block  209 B, the digital equalizer  106 B will have a natural tendency to revert to “ideal” compensation of the channel frequency response, since the feedback loop will then involve only the DFE  107 B, which is empowered with the capability to reduce ISI. In this case, when a new set of values for the coefficients  212  is reached, i.e., after a new settling time, the power of the second error signal  231 B at convergence will actually increase, i.e., the data decisions made by the DFE will be less accurate than they were when consideration had been given to error signal  263 B. 
     Therefore, it is important to program the control block  214  to weight the error signals  263 B and  231 B according to an appropriate ratio. The ideal value for this ratio will depend on many factors, such as the amount of crosstalk present and the degree to which the offset filter  208 B correctly represents the crosstalk frequency characteristic. 
     In summary, the first alternate embodiment of the present invention just described achieves a substantial reduction in crosstalk noise at the expense of injecting a tolerable amount of inter-symbol interference at the input to the DFEs  107 A,B. The desired amount of crosstalk noise attenuation as a function of frequency is modifiable via the coefficients  240  of the offset filter  208 B. At the same time, variations in the channel are automatically tracked by the algorithms running in the control blocks  214 ,  227  of the digital equalizer  106 B and DFE  107 B, respectively. 
     In yet another refinement of the present invention, the coefficients of the offset filter need not be fixed, but may be dynamically adjusted so as to exhibit continuously improving crosstalk cancellation behaviour. In FIG. 3, illustrating a second alternate embodiment of the present invention, the second error signal  231 B output by the comparator  216  to the control block  227  in the DFE  107 B is also fed to a control block  342  in the offset filter  208 B for controlling the offset filter coefficients  240 . 
     The embodiment of FIG. 3 will function similarly to that of FIGS. 2A and 2B, but will undergo an additional step, namely, adaptation of the coefficients  240  of the offset filter  208 B based on the second error signal  231 B supplied by the DFE  107 B. Upon convergence of the offset filter coefficients  240 , the frequency response of the offset filter  208 B provides a more precise estimate of the crosstalk coupling characteristic, which was until this point held fixed to resemble a somewhat upwards-sloping curve. 
     While it is acceptable to allow the control block  342  to freely adapt at the offset filter coefficients  240 , thereby leading to improved performance, it is still preferable to provide certain bounds within which the shape of the offset filter frequency response must fall. This is to avoid the scenario in which the offset filter coefficients adapt to a point beyond which the frequency response provided no longer resembles the crosstalk frequency characteristic. 
     Having regard now to the present invention in general, it is to be considered that, from a practical point of view, the digital filtering, summation and comparison operations may be performed by one or more digital signal processors or general purpose microprocessors. Furthermore, analog-to-digital conversion, while necessary at some point prior to decision feedback equalization, may be performed at any stage, for example, subsequent to demodulation or even at the outset, prior to bandpass filtering. In the latter case, any “analog” filtering or equalization operations would be replaced by digital ones. 
     Also, it is possible to envisage another embodiment of the present invention in which the analog equalizer is adapted in a manner similar to the way in which the coefficients of the digital equalizers are controlled in the first and second alternate embodiments. Of importance is only the combined frequency response of both equalizers, taking into account the operating band of each filter, i.e., passband for analog filtering and baseband for digital equalization. Alternatively, analog equalization may be omitted, or digital equalization may be omitted, in which case the feedback loop from the error calculation block  209 B can be made to control modification of the parameters of the analog filter via a control block on a microprocessor. 
     In view of the above description of the preferred and alternate embodiments of the present invention and the numerous possible variations thereof, the scope of the invention is only to be limited by the claims appended hereto.