Abstract:
A dual-supply line-interface circuit ( 100 ) uses a −48V power supply (V BAT1 ) to drive long subscriber loops ( 120 ) and uses a −28V power supply (V BAT2 ) to drive short subscriber loops. For intermediate-length loops, a dual-slope current-feed profile (FIG.  4 ) is employed to limit the line-circuit&#39;s power dissipation. The line-interface circuit operates in an apparent constant-current mode, generating about 40 mA of differential line current using the low power supply, up to a threshold line voltage of about 25V, which is equal to the low power supply voltage minus required overhead. For longer loops, the line-interface circuit switches to a second constant-current mode, generating about 22 mA of differential current using the high power supply, which maintains the loop current constant until it drops to the 48V resistive-feed value.

Description:
TECHNICAL FIELD 
     This invention relates generally to analog telephone line interface circuits, and specifically to the battery feed circuits of such line interface circuit. 
     BACKGROUND OF THE INVENTION 
     Conventional analog telephone line-interface circuits, also known as analog port circuits, require a 48VDC power supply for operation and for reliable signaling on long subscriber loops (telephone lines). Long loops have a high resistance relative to short loops, and therefore require a relatively high voltage to drive them. The circuit which couples the DC power to the telephone line is known as a battery-feed circuit. Even though battery-feed circuits commonly employ current-limiting and limit loop current to 42mA, 2W of power can be dissipated by the line-interface circuit. This high power dissipation limits the number of line-interface circuits that can be integrated on a single integrated-circuit device (a “chip”), as well as the number of telephone lines that can be served by a single 48V power supply. 
     To reduce power dissipation, the art has employed dual-supply line-interface circuits. These circuits employ a second power supply having a voltage lower than the high-voltage (48V) power supply, for powering short subscriber loops. 
     SUMMARY OF THE INVENTION 
     In order to reduce even further the power dissipated by a dual-supply line-interface circuit, a dual-slope current-limit profile is employed for operation of the line-interface circuit to effect current limiting. The second power supply preferably operates at 28V, which can be generated from the high-voltage (48V) supply via a DC-to-DC converter. This significantly increases the supply current that is made available by the line-interface circuit to short subscriber loops, and thus significantly increases the number of short subscriber loops which the power supply can handle. For example, assuming 90% efficiency of the converter, the supply current and the short-loop-handling capacity of the power supply are increased by 50%. The 48V supply is still used directly to drive long loops. For intermediate-length loops, the dual-slope current-feed profile is employed to limit the line-interface circuit&#39;s power dissipation. The line-interface circuit operates in an apparent constant-current mode using the low power supply up to a threshold line voltage which is equal to the low power supply voltage minus required overhead. For longer loops, the line-interface circuit switches to a second constant-current mode which is substantially lower than the constant current for the shorter loops, which maintains the loop current constant until the loop current drops to the 48V resistive-feed value (the minimum value required to drive a telephony device connected to the loop). 
     Generally according to the invention, a line-interface circuit for connecting to an analog telephone line that comprises a pair of leads (e.g., tip and ring leads) has a battery-feed circuit that monitors line voltage across the pair of leads and substantially maintains line current flowing between the leads at one of two substantially constant values. When the line voltage is exceeded by a first threshold voltage (e.g., ˜25V), the battery-feed circuit maintains the line current at a first substantially-constant value (e.g., 40 mA). When the line voltage exceeds a second threshold voltage (e.g., ˜25.5V), the battery-feed circuit maintains the line current at a second substantially-constant value (e.g., 22 mA). If the two thresholds are not one and the same, the battery-feed circuit preferably varies the line current between the first and the second values as the line voltage varies between the first and the second thresholds. Preferably, the line current monitored by the battery-feed circuit is differential current between the two leads. More specifically according to a preferred embodiment of the invention, the battery-feed circuit comprises a driver for driving (powering) the line which uses a first power supply of dual power supplies to drive the line while the line current is at the first current value, and uses a second power supply of the dual power supplies to drive the line while the line current is at the second value. The dual power supplies operate at voltages of significantly different magnitudes—for example, the first power supply operates at −28VDC and the second power supply operates at −48VDC. 
     Illustratively, the battery-feed circuit includes a current-feedback loop that includes a constant-current supply that generates a constant current for driving the feedback loop to produce a constant current of one of the first and the second current values on the line. The feedback loop further includes a variable-current supply that generates a variable current that combines with the constant current generated by the constant-current supply to drive the feedback loop. The variable current varies with the line voltage to cause the feedback loop to produce the constant current of the one current value on the line when the line voltage is exceeded by the first threshold value, and to cause the feedback loop to produce a constant current of another of the first and second current values on the line when the line voltage exceeds the second threshold value. The variable current further illustratively causes the feedback loop to produce a line current that varies between the first and the second current values as the line voltage varies between the first and the second threshold values, and vice versa. 
     In one implementation, a line-interface circuit for connecting to an analog phone line comprising a pair of leads has a battery-feed circuit that powers the line from one of a pair of power supplies operating at significantly different voltages. The battery-feed circuit comprises a pair of drivers, each driving a different one of the pair of leads and each sensing voltage on the different one of the pair of leads. One driver uses a first one of the pair of power supplies to drive the line while the differential current on the leads of the line is at a first value, and uses a second one of the pair of power supplies to drive the line while the differential current is at a second value. The two power supplies operate at voltages of significantly different magnitude. The battery-feed circuit also includes a differential-current sensor for sensing the differential current flowing between the pair of leads and generating a first voltage representative of the differential current. The first voltage is used to control a second voltage at a junction. The battery-feed circuit further includes a transconductance amplifier that drives the one of the pair of drivers. It has an input connected to the junction. A variable-current source generates a variable current at the junction as a function of line voltage in order to create a variable said second voltage at the junction. The net effect is that the differential-current sensor, the variable-current generator, the transconductance amplifier, and the one driver form a current-feedback loop that maintains the differential current at a substantially constant first value when the line voltage is below the first threshold value, and maintains the differential current at a substantially constant second value significantly smaller than the first value when the line voltage is above the second threshold value, greater than the first threshold value. 
     These and other advantages and features of the invention will become more apparent from the following description of an illustrative embodiment of the invention considered together with the drawing. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     FIG. 1 is a partial circuit-and-block diagram of a telephone line-interface circuit that embodies an illustrative example of the invention; 
     FIG. 2 is a partial circuit diagram of a variable-current supply of the telephone line-interface circuit of FIG. 1; 
     FIG. 3 is a diagram of the operational characteristic of the variable-current supply of FIG. 2; 
     FIG. 4 is a diagram of the operational characteristic of the telephone line-interface circuit of FIG. 1; and 
     FIG. 5 is a circuit diagram of an amplifier of the telephone line-interface circuit of FIG.  1 . 
    
    
     DETAILED DESCRIPTION 
     FIG. 1 shows those portions of a telephone line-interface circuit  100  that are relevant to an understanding of this invention. Circuit  100  is illustratively an L7500-series or an L8500-series subscriberline-interface circuit (SLIC) integrated-circuit device of Lucent Technologies Inc. The SLIC utilizes a voltage-feed current-sense architecture, wherein a pair of voltage sources feed the DC power as well as the voice-band signal to a telephone line  120 , and the signal from the far end (e.g., a telephone) is sensed by a differential-current-sense circuit that is connected in series with line  120 . The impedances which the SLIC presents to line  120  can be synthesized by the gain around the feedback loop. 
     Circuit  100  includes a pair of amplifiers AT  103  and AR  104  that are connected through a differential-current sensor  105  to the tip lead  101  and the ring lead  102 , respectively, of telephone line  120  and deliver current thereto. The delivered current enables the telephone switching system to detect the presence and status of equipment (e.g., a telephone) connected to telephone line  120 . Circuit  100  also couples audio signals from line  9  to telephone line  120  and from telephone line  120  to line L 1 . 
     Power amplifiers  103  and  104  are voltage-mode operational amplifiers operating in unity-gain configuration to transmit onto line  120  audio signals supplied to their positive inputs by transmit line L 2  through a level-shift circuit  123 . Tip lead  101  provides negative feedback to amplifier AT  103 , while ring lead  102  provides negative feedback to amplifier AR  104 . The positive input of amplifier  103  is connected through an impedance-matching buffer  115  to a voltage source V CF1 , which in this example provides approximately ˜2 VDC. The positive input of amplifier  104  is connected through an impedance-matching buffer  116  to a voltage V CF2 . V CF2  is produced by forcing a current generated by a current supply  125  into a resistor  114  that is connected to the V BAT1  (−48 VDC) supply rail. Illustratively, the current output by current supply  125  is 50 μA and resistor  114  is 100 kΩ, so V CF2  is −43 VDC (−48V+50 μA*100 kΩ) when the loop current in line  120  is zero. Amplifiers  103  and  104  supply V CF1  and V CF2  to tip and ring leads  101  and  102 , respectively. 
     Differential current sensor  105  detects the difference in current flowing on leads  101  and  102  and puts out an indication of that difference to a negative input of an amplifier AX  106 . A positive input of amplifier  106  is connected to ground. Amplifier  106  amplifies the difference indication by a magnitude determined by a feedback resistor  107  which connects the output V ITR  of amplifier  106  back to the negative input of amplifier  106 . In this illustrative example, with no loop current flowing in line  120 , output V ITR  of amplifier  106  is at 0V. With loop current flowing in line  120  in the normal direction (from tip lead  101  to ring lead  102 ), output V ITR  of amplifier  106  is negative. The transimpedance gain from the differential loop current to V ITR  is about 250V per one Ampere of differential current. The output V ITR  of amplifier  106  drives signal line V ITR    121 . Line V ITR    121  is connected to audio receive line L 1  through a DC-blocking capacitor  122 . Line V ITR    121  is also connected through a current-limiting resistor  108  to a junction  124  with the output of a current supply  109 . Current supply  109  is connected to the supply rail V CC , which in this example is +5 VDC, and outputs a constant current of 75 μA to junction  124  in this example. 
     Junction  124  is connected to a transconductance stage  111 - 113  which includes an operational amplifier  111 , a PNP transistor  112 , and a resistor  113 . Junction  124  is connected to a positive input of operational amplifier  111 . The output of operational amplifier  111  is connected to the base of transistor  112 . The emitter of transistor  112  is connected to the negative input of operational amplifier  111 , and through resistor  113  to ground. The collector of transistor  112  is connected to V CF2 . If the voltage at junction  124  is positive, then the current output from the collector of transistor  112  is zero. However, if the voltage at junction  124  is negative, then the current output from the collector of transistor  112  is equal to the voltage at junction  124  divided by resistor  113 . The current from the collector of transistor  112  is fed into resistor  114  and therethrough to V BAT1 . The voltage gain from junction  124  to V CF2  is inverting (a gain of −50 in this example) for junction  124  having negative voltages. For junction  124  having a voltage of zero or a positive voltage, the gain is zero. The transimpedance gain from the loop current of line  120  to V ITR    121  is 250 V/A, as stated earlier. Then the input impedance which circuit  100  presents to line  120  is 12.5 kΩ (250 V/A*50). This is the impedance value when circuit  100  is in loop-current-limiting mode. 
     The voltage at junction  124  is determined by the voltage on line V ITR    121 , resistor  108 , and current supply  109 . As stated earlier, line V ITR    121  is at 0V when the loop current is at zero; hence, the voltage at junction  124  is positive. As the loop current flows, as stated earlier, voltage on line V ITR    121  becomes negative. The loop current for which junction  124  becomes 0 VDC is the current limit for the SLIC. 
     As described so far, line circuit  100  is conventional. According to the invention, however, by varying the current supplied to junction  124 , the current limit of circuit  100  can be changed. Junction  124  is also connected to the input of a second current supply  110 . Current supply  110  is driven by a voltage V BAT1 , which in this example is −48 VDC, and sinks a variable current I PROG  from junction  124 , which in this example varies from 0 to 34 μA. Hence, the net current at junction  124  is a variable current of 41 to 75 μA. The amount of current sinked by current supply  110  is a function of the difference between a voltage V BAT2 , which in this example is −28 VDC, and V CF2 . Both of these voltages are connected to current supply  110 . 
     FIG. 2 shows the structure of relevant parts of variable current supply  110 . An NPN transistor  200  has its collector connected to junction  124 , its base connected through a voltage supply  220  to V BAT2 , and its emitter connected to the base of a second NPN transistor  201 . Voltage supply  220  keeps the base of transistor  200  at about 2.8 VDC above V BAT2 . The collector of transistor  201  is connected to junction  124 , and its emitter is connected to an input of a diode  203 . A resistor  202  connects the base of transistor  201  to its emitter. Together, transistors  200  and  201  and resistor  202  form a Darlington pair. 
     In a symmetrical configuration, an NPN transistor  210  has its collector connected to ground, its base connected to V CF2 , and its emitter connected to the base of a second NPN transistor  211 . The collector of transistor  211  is connected to ground, and its emitter is connected to an input of a diode  213 . A resistor  212  connects the base of transistor  211  to its emitter. Together, transistors  210  and  211  and resistor  212  also form a Darlington pair. 
     The outputs of diodes  203  and  213  are respectively connected to the collectors of NPN transistors  205  and  207 , and are interconnected by a resistor  204 . The bases of transistors  205  and  207  are connected to a biasing voltage source V NR1 , which is adjusted to cause each transistor  205  and  207  to draw 17 μA of current. The emitters of transistors  205  and  207  are respectively connected across resistors  206  and  208  to V BAT1 . 
     The operation of variable current supply  110  is as follows. When V CF2 −V BAT2  is less than 2.8V—the voltage at the base of transistor  200 —transistors  210  and  211  are turned off and transistors  200  and  201  are turned on and conducting the 34 μA that are being drawn by transistors  205  and  207  away from junction  124 , thereby resulting in 41 μA of current across resistor  108 . When V CF2 −V BAT2  is more than the 2.8V at the base of transistor  200 , transistors  200  and  201  are turned off and not conducting current from junction  124  while transistors  210  and  211  are turned on and conducting from ground (and not from junction  124 ) the 34 μA that are being drawn by transistors  205  and  207 . This results in the full 75 μA of current output by current source  109  across resistor  108 . When V CF2 −V BAT2  is substantially at 2.8V, transistors  200  and  201  and  210  and  211  are partially on, resulting in a narrow transition region where between 0 and 34 μA are being conducted by current source  110  away from junction  124 . 
     The operational characteristic of current supply  110  is shown in FIG.  3 . While the voltage difference V CF2 −V BAT2  is below a first threshold of about 2.5V, supply  110  sinks 34 μA of current. Above this threshold in the vicinity of 2.8V, supply  110  sinks current in proportion to the voltage difference, up to a second threshold of about 3.1 V, at which point supply  110  sinks no current. Beyond the second threshold, supply  110  continues to sink no current. 
     The resulting current-limiting operation of line circuit  100  of FIG. 1 is as shown in FIG.  4  and described below. While line  120  is not in use, the voltage V TR  between tip lead  101  and ring lead  102  (where V TR =V CF2 =V CF1 ) is about 41V, the current I TR  from tip lead  101  to ring lead  102  is zero, the differential current on leads  101  and  102  of telephone line  120  is also zero, so the voltage on V ITR  line  121  is 0, and the current produced by current supplies  109  and  110  at junction  124  is 41 μA (i.e., 75 μA−34 μA), which produces a 5V drop across resistor  108 , i.e., a 5V level at junction  124 , thereby turning off high-gain stage cascade  111 - 113 . With cascade  111 - 113  turned off, current supply  125  and resistor  114  keep V CF2  at about −43V. This produces a difference of about −15V between V CF2  and V BAT2 , which (see FIG. 3) causes current generator  110  to sink 34 μA of current from junction  124 . 
     When line  120  comes into use (e.g., a telephone goes “off hook” on line  120 ) V TR  begins to drop, and when it drops to about 41V, loop current begins to flow in line  120 . The loop current in line  120  increases to about 22 mA as V TR  drops to about 39.5V. At this point, line V ITR    121  is sufficiently negative so that junction  124  is at 0VDC (41 μA*133 kΩ/250), high-gain cascade  111 - 113  turns on and limits the loop current in line 120 to about 22 mA as V TR  drops further. When VTR drops to about 25.5V, I PROG  current output by circuit  110  starts to decrease from 34 μA to zero. The net current flow output of junction  124  to resistor  108  is increased from 41 μA to 75 μA as V TR  drops further to 24.9V. Any further decrease in V TR  does not result in increased current output from junction  124  into resistor  108 ; therefore, the loop current in line  120  stays at a relatively constant value of about 40 mA. 
     In order to take full advantage of this DC feed profile for power-feeding efficiency, amplifier AR  104  must be modified from its traditional three-stage configuration. FIG. 5 shows such a simplified voltage-mode operational amplifier. Essentially, the modification involves adding a fourth stage comprising a current-steering transistor and a diode to the amplifier output. The first stage, comprising a current source  500  and transistors  502 - 505 , is a transconductance amplifier, which outputs a current at junction  508  into the base of a transistor  506 . The second stage, comprising a current source  501  and the transistor  506 , is a common-emitter amplifier, which takes the output current from the first stage and beta-multiplies it to its collector output, junction  509 . A Miller capacitor  507  connected between junctions  508  and  509  compensates the operational amplifier to ensure stable unity gain. The third stage is a push-pull amplifier, comprising transistors  510  and  511 , which provides the drive capability to the output load. In order to take advantage of V BAT2  being a lower supply voltage than V BAT1 , a current-steering transistor  512  is incorporated in the design. It works in the following manner. If V out −V BAT2  is greater than 2.5V, transistor  512  is in its active mode, and the load current sink from junction  509  flows to V BAT2  through a diode  513 . 
     Only a small fraction of current (1/(1+beta)) of the load current flows into the emitter of transistor  511  and to V BAT1 . If V OUT −V BAT2  is less than 2.5V, transistor  512  is in saturation and cannot support the load current with high beta; the load flows through the base-emitter junction of transistor  512  into the emitter of transistor  511  and to V BAT1 . The threshold of 2.5V is controlled by the forward-on voltage of diode  513  and the internal collector resistance of transistor  512  times the worst-case loop current. This 2.5V threshold is also incorporated into the design of circuit  110  to ensure that, when the load current is steered from V BAT2  to V BAT1 , the tip and ring current limit has already reached 22 mA, thereby minimizing the SLIC chip internal power dissipation. 
     Of course, various changes and modifications to the illustrative embodiment described above will be apparent to those skilled in the art. For example, the circuitry can be implemented from active components having an opposite polarity to that shown. Also, the circuitry can be implemented using different circuit technologies or circuit designs. Such changes and modifications can be made without departing from the spirit and the scope of the invention and without diminishing its attendant advantages. It is therefore intended that such changes and modifications be covered by the following claims.