Abstract:
A switched capacitor integrator particularly suitable to realize low-pass filters without inducing noise on the nodes of the reference potentials of the integrator, is provided by halving the input capacitance during an operating phase, and by transferring the electric charge between the input switched capacitance and the capacitor of integration of one and the other feedback branch of the differential amplifier, in a direct manner, that is, not referred to a fixed common potential. A unique current path is established, thus averting the effects caused by inevitable mismatches between the integrated capacitors.

Description:
FIELD OF THE INVENTION 
     The present invention relates to interfaces, and, more particularly, to Digital/Analog (D/A) and Analog/Digital (A/D) conversions. 
     BACKGROUND OF THE INVENTION 
     D/A and A/D conversions are important interface processes that find applications in telecommunication, audio systems, instrumentation and the like. These processes continuously demand highly precise filters, such as the switched capacitor filters whose basic element is the integrator. Therefore, a particularly efficient integrator may be very useful for the above-cited applications. 
     By way of example, consider the application of a sigma-delta (ΣΔ)D/A converter for low-pass signals whose basic scheme is as shown in FIG.  1 . In these types of converters, the ΣΔ modulator converts an N bit digital signal into a 1-bit signal, shifting the quantization noise thus introduced outside the signal band. The low-pass analog filter would then eliminate this noise. The purpose of a 1-bit DAC is to form the interface between the digital world, represented by the bit-stream output delivered by the modulator, and the analog world, represented by the switched capacitor low-pass filter. 
     FIG. 2 shows the output of the 1-bit DAC in detail. Specifically, FIG. 2 shows how the energy that corresponds to the logic level 1 is different from the one that corresponds to the level 0. The figure also shows how the energy of two consecutive  1 s is different from the energy that may be obtained by adding two nonconsecutive 1s. For these reasons, switched capacitor filters are preferred because what is considered important is the value assumed by the signal of FIG. 2 at a specific instant. 
     FIG. 2 also shows the effect in time of unwanted noise, as generated by the voltage references V h  V l , that becomes superimposed on the useful signal. In frequency, the high frequency noise contained in the input signal is eventually folded back in the band, thus determining a notable degradation of the total signal/noise ratio, as shown in FIG.  3 . The effect of folding back the noise inside the band is even more evident as the noise that superimposes the signal, delivered as output by the 1-bit DAC, depends on the bit-stream. Indeed, the expression for the filter&#39;s output voltage is given by:                V   OUT     =         V   ref     ·     K       2   N     -   1       ·     α        (   K   )         +   C             (   1   )                                
     where: 
     K:input word to the DAC (O≦K≦2 N −1); 
     N:number of bits of the converter; 
     C:constant that allow the curve to be translated; 
     α(K):variable, K dependent, which expresses the perfect linearity of the converter; 
     V ref ∝V h -V l : reference voltage. 
     Equation 1 demonstrates that the existence of a nonlinearity introduces a multiplying factor of V ref , thus causing a distortion effect. 
     In the case of a 1 bit converter (N=1) shown in FIG. 1, the system is always linear (α(K)=1∀K) as long as V ref  is perfectly stable and free of components whose frequency is different from 0. If this were not the case, such components would modulate the spectrum components of the input signal, thus determining a distortion effect. 
     Hence, there is a need for an efficient switched capacitor integrator structure, which permits the realization of the desired filters without introducing excessive and undesired noises on the reference voltages V h  e V l . 
     The 1-bit DAC is often directly integrated inside the following switch capacitor filter, especially for those applications to be integrated on a single chip. Since the integrator is usually the basic element of a filter, there exist different structures that permit the 1-bit DAC to link to the integrator itself. Depicted in FIG. 4 is a conventional structure of a switched capacitor integrator with a stray-insensitive connection of the input capacitance C 11 . 
     FIG. 5 shows a known evolution of the integrator of FIG. 4, which doubles the dynamic of the output signal by using a fully differential structure. The capacitances C a , C b , C c  and C d  represent the parasitic capacitances of the capacitors C 1  and C 11 , realized on the silicon. Typically, one of these two parasitic capacitances is much bigger than the other because the distance between one of the two plates and the substrate connected to ground is shorter. 
     By putting V din =2·(V h -V l ), we have:                  V   dout          (   n   )       =         V   dout          (     n   -   1     )       -           C   11     +       C   b     /   2         C   1       ·       V   din          (     n   -   1     )       ·       (     -   1     )     K                 (   2   )                                
     where K=1 with bs=1 and K=0 with bs=0 (bs is the bit-stream). 
     Equation 2 is obtained upon considering the components of the circuit of FIG. 5 as ideal components, with the exception of the capacitors to which are associated parasitic capacitances. In addition, the operational amplifier should be able to maintain the mean value of the outputs to a constant value in time. Furthermore, by indicating the asymptotic voltage on the input nodes of the operational amplifier as V x (∞) we have:                  V   x          (   ∞   )       =         2   ·     C   11     ·     V   com       +       (       V   h     +     V   1       )     ·     C   b           2   ·     (       C   11     +     C   b       )                 (   3   )                                
     Equation 2 and Equation 3 emphasize the need to minimize the capacitance C b , while the other parasitic capacitances are not involved, with the exception of C c  which contributes in determining the time required by the system to obtain V x =V x (∞). 
     The fundamental problem of these approaches is that the capacitive load on the reference voltages V h  e V l  depends on the bit-stream; a condition that may introduce undesired distortion effects. 
     The scheme of FIG. 6 was developed to avert this problem. This approach permits a constant load on the voltages V h  and V l , regardless of the bit-stream value. The functioning characteristics of this modified scheme are identical to the ones shown in the scheme of FIG.  5 . 
     SUMMARY OF THE INVENTION 
     A switched capacitance integrator has now been devised which is particularly more efficient and suitable to realize filters without introducing significant noises on the nodes of the reference potentials of the integrator. 
     As compared to an earlier approach, e.g. identifiable in a scheme as the one shown in FIG. 6, the fundamental aspect of the present invention includes halving the whole input capacitance to be charged during an operating phase, and in carrying out the transfer of electric charge between the input switched capacitor and the capacitor of integration of one and the other feedback branch of the operational amplifier in a direct manner, that is, not referred to a common fixed potential as it is the case of known circuits. This reduces the strain on the operational a amplifier because the current delivered through an output node in order to discharge a capacitor is identical to the current delivered through the other output node that charges the other capacitor. Hence, there exists a unique current path, thus averting the effects caused by eventual mismatches between capacitances, unlike what happen in the known circuits of FIGS. 5 and 6 in which the current path involves all four capacitors, that is, the pair of input capacitors and the pair of capacitors of integration, thus increasing the problems generated by the capacitive mismatch. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1, as already mentioned is a basic scheme of a D-A D converter. 
     FIG. 2 shows the form of the output signal of the 1-bit DAC converter. 
     FIG. 3 shows the spectrum that the converter delivers as output in the presence of noises on the nodes of the reference potentials, dependent on the bit-stream. 
     FIG. 4 shows the conventional scheme of a switched capacitor integrator using a 1-bit DAC converter. 
     FIG. 5 shows a known embodiment of a fully differential integrator using a 1-bit DAC converter. 
     FIG. 6 shows another known structure of a differential integrator with a 1-bit DAC differential integrator. 
     FIG. 7 shows the structure of a fully differential integrator with a 1-bit DAC in accordance with the present invention. 
     FIG. 8 a  and FIG. 8 b  compare the functional scheme of a known structure with the functional scheme of the structure of the invention, during the phase of integration. 
     FIG. 9 shows the functioning diagrams of a known structure of FIG. 5 as well as a structure of the invention of FIG. 7 intended for three different functioning conditions. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     As in the example of the known structure of FIG. 6, the structure of the invention illustrated in FIG. 7 allows for a load on the two nodes of the reference voltages V h  and V l  regardless of the bit-stream. The ensuing description outlines the further advantages of the present invention. 
     During the phase 1, the input capacitance C 11  of the circuit of the invention shown in FIG. 7 is charged. C 11  is realized in an integrated form through two capacitors whose capacitance value is halved C 11 /2, preferably with plates that are directly opposite each other in order to have equal parasitic capacitances on V h  and V l . The input capacitance C 11  is discharged during the successive phase through the two virtual grounds of the operational amplifier, that is, the inverting input (−) and the noninverting input (+). 
     As compared to the known circuits of FIGS. 5 and 6, the first advantage includes halving the total capacitance to be charged during the phase 1. Indeed, in FIGS. 5 and 6, two C 11  capacitances should be charged whereas the circuit of the invention requires the charge of two C 11 /2 capacitances. This condition reduces the strain on the nodes of the two reference voltages V h  and V l , thus reducing the probability of introducing distortion effects. Moreover, the decrease of the integrated capacitance results in a non-negligible reduction of silicon area. 
     Unlike the two known schemes of FIGS. 5 and 6, the transfer of the electric charge between the capacitances C 11  and C 1  occurs in a direct manner rather than through the fixed common potential V com . In substance, the charge removed from the input capacitance C 11  is absorbed by one of the capacitors of integration C 1  and thereafter transferred on the other one through a unique current path. This lessens the strain on the operational amplifier because the current supplied by an output clamp in order to discharge the capacitance of integration C 1 , is identical to the current that charges the other capacitance of integration C 1  through the other output clamp. As a result, there exists a unique path C 1 C 11 C 1  through which the charge flows, thus limiting the effects caused by any mismatch that may exist among the capacitors. 
     Instead, in the cases shown in FIG.  5  and FIG. 6 the charging path involves all four capacitances, which is a condition that increases the problems caused by the additional mismatch of the two input capacitors C 11 . 
     A further advantage of the approach of the present invention is the augmented speed because the feedback moves toward opposite directions both input potentials of the operational amplifier (immediately after the switches of phase 2 are closed), thus discharging more rapidly the double capacitance C 11 /2. 
     Further clarification of the present invention will be discussed while referring to FIGS. 8 a  and  8   b  which are relative to the instant that follows the start of the second phase of integration. These figures compare the known system (FIG. 8 a ) with the scheme of the invention (FIG. 8 b ). Under the hypothesis that an ideal components situation exists, with the exception of the capacitors, the fundamental relationships that rule the scheme of FIG. 7 are:                  V   dout          (   n   )       =         V   dout          (     n   -   1     )       +           -     V   din       ·     C   11     ·       (     -   1     )     K       +         V   X          (   n   )       ·     (       C   b1     -     C   b2       )       +     f        (   K   )         C1               (   4   )                                
     where                f        (   K   )       =                {         -     C   b2       ·     V   l       +       C   b1     ·     V   h                                  if                 K     =   1                              ⌊         -     C   b2       ·     V   h       +       C   b1     ·     V   1                                  if                 K     =   0                                  
     From Equation 4 the importance of C b1 =C b2  is evident. 
     For this reason, the capacitance C 11  of FIG. 7 is obtained with two integrated capacitors C 11 /2 crossing one another. In this way C b1  and Cb 2  would be certainly identical. However, unlike the other two schemes considered, it is not possible in this case to connect the smaller parasitic capacitance to the inputs of the operational amplifier in order to minimize its effect, because there will be two identical parasitic capacitances, equal to the sum of the two parasitic capacitances of the respective C 11 /2. This disadvantage is less than what it may seem because the capacitances are halved and the parasitic capacitances are also reduced. 
     By considering a common mode condition in which the output is kept at a value V com , and by indicating the parasitic capacitances as {overscore (C)} b =C b1 =C b2  and upon substituting in Equation 4, we have:                  V   dout          (   n   )       =         V   dout          (     n   -   1     )       -           C   11     +         C   _     b     /   2         C   1       ·     V   din     ·       (     -   1     )     k                 (   5   )                                
     In addition, we also have:                  V   x          (   ∞   )       =         V   h     +     V   1       2             (   6   )                                
     From equation 6 it may be noticed that none of the capacitances has any role in determining the asymptotic value of the inputs of the operational amplifier, which is purely fixed by the reference voltages V h  and V l . 
     By analyzing the effect of an eventual unwanted noise existing on the reference voltages V h , V l  and V com , the three circuits being analyzed do not present any differences. By way of example, the circuits described in FIG.  5  and FIG. 7 have been simulated in order to test the above description. The results are shown in FIG. 9 in which the common mode of the output of the circuit shown in FIG. 7 is faster than the one of FIG.  5 . 
     In addition, the differential signal of the circuit of the invention reaches the steady state value in a shorter time span than the other schemes considered. FIG. 9 also highlights that in the circuit of the invention there is a reduction of the current supplied by the operational, and the inputs of the operational amplifier are less perturbed.