Abstract:
A microwave ablation system includes a generator operable to output energy and an antenna coupled to the generator via a coaxial cable. The antenna delivers energy to tissue and includes a proximal radiating section and a distal radiating section. The antenna also includes a feedpoint section defined by the portion of the distal radiating section that underlaps the proximal radiating section.

Description:
BACKGROUND 
       [0001]    1. Technical Field 
         [0002]    The present disclosure relates generally to microwave ablation procedures that utilize microwave surgical devices having a microwave antenna which may be inserted directly into tissue for diagnosis and treatment of diseases. More particularly, the present disclosure is directed to an integrated matching network in the feedpoint structure of a microwave dipole antenna. 
         [0003]    2. Background of Related Art 
         [0004]    In the treatment of diseases such as cancer, certain types of cancer cells have been found to denature at elevated temperatures (which are slightly lower than temperatures normally injurious to healthy cells.) These types of treatments, known generally as hyperthermia therapy, typically utilize electromagnetic radiation to heat diseased cells to temperatures above 41° C., while maintaining adjacent healthy cells at lower temperatures where irreversible cell destruction will not occur. Other procedures utilizing electromagnetic radiation to heat tissue also include ablation and coagulation of the tissue. Such microwave ablation procedures, e.g., such as those performed for menorrhagia, are typically done to ablate and coagulate the targeted tissue to denature or kill the tissue. Many procedures and types of devices utilizing electromagnetic radiation therapy are known in the art. Such microwave therapy is typically used in the treatment of tissue and organs such as the prostate, heart, liver, lung, kidney, and breast. 
         [0005]    One non-invasive procedure generally involves the treatment of tissue (e.g., a tumor) underlying the skin via the use of microwave energy. The microwave energy is able to non-invasively penetrate the skin to reach the underlying tissue. However, this non-invasive procedure may result in the unwanted heating of healthy tissue. Thus, the non-invasive use of microwave energy requires a great deal of control. 
         [0006]    Presently, there are several types of microwave probes in use, e.g., monopole, dipole, and helical. One type is a monopole antenna probe, which consists of a single, elongated microwave conductor exposed at the end of the probe. The probe is typically surrounded by a dielectric sleeve. The second type of microwave probe commonly used is a dipole antenna, which consists of a coaxial construction having an inner conductor and an outer conductor with a dielectric junction separating a portion of the inner conductor. The inner conductor may be coupled to a portion corresponding to a first dipole radiating portion, and a portion of the outer conductor may be coupled to a second dipole radiating portion. The dipole radiating portions may be configured such that one radiating portion is located proximally of the dielectric junction, and the other portion is located distally of the dielectric junction. In the monopole and dipole antenna probe, microwave energy generally radiates perpendicularly from the axis of the conductor. 
         [0007]    The typical microwave antenna has a long, thin inner conductor that extends along the axis of the probe and is surrounded by a dielectric material and is further surrounded by an outer conductor around the dielectric material such that the outer conductor also extends along the axis of the probe. In another variation of the probe that provides for effective outward radiation of energy or heating, a portion or portions of the outer conductor can be selectively removed. This type of construction is typically referred to as a “leaky waveguide” or “leaky coaxial” antenna. Another variation on the microwave probe involves having the tip formed in a uniform spiral pattern, such as a helix, to provide the necessary configuration for effective radiation. This variation can be used to direct energy in a particular direction, e.g., perpendicular to the axis, in a forward direction (i.e., towards the distal end of the antenna), or combinations thereof. 
         [0008]    Invasive procedures and devices have been developed in which a microwave antenna probe may be either inserted directly into a point of treatment via a normal body orifice or percutaneously inserted. Such invasive procedures and devices potentially provide better temperature control of the tissue being treated. Because of the small difference between the temperature required for denaturing malignant cells and the temperature injurious to healthy cells, a known heating pattern and predictable temperature control is important so that heating is confined to the tissue to be treated. For instance, hyperthermia treatment at the threshold temperature of about 41.5° C. generally has little effect on most malignant growth of cells. However, at slightly elevated temperatures above the approximate range of 43° C. to 45° C., thermal damage to most types of normal cells is routinely observed. Accordingly, great care must be taken not to exceed these temperatures in healthy tissue. 
         [0009]    In the case of tissue ablation, a high frequency electrical current in the range of about 500 mHz to about 10 gHz is applied to a targeted tissue site to create an ablation volume, which may have a particular size and shape. Ablation volume is correlated to antenna design, antenna performance, antenna impedance and tissue impedance. The particular type of tissue ablation procedure may dictate a particular ablation volume in order to achieve a desired surgical outcome. By way of example, and without limitation, a spinal ablation procedure may call for a longer, narrower ablation volume, whereas in a prostate ablation procedure, a more spherical ablation volume may be required. 
         [0010]    In microwave ablation devices, when microwave energy is delivered to an antenna by a feed line, the impedance of the antenna and feed line must match exactly for maximum energy transfer from the feed line to the antenna to be possible. The impedance of the antenna varies based on many factors including the antenna&#39;s natural resonance at the frequency being transmitted and the size of the conductors used to construct the antenna. When an antenna and feed line do not have matching impedances, some of the electrical energy cannot be transferred from the feed line to the antenna. Energy not transferred to the antenna is reflected back towards the transmitter. It is the interaction of these reflected waves with forward waves which causes standing wave patterns. Reflected power has three main implications: microwave frequency (MW) energy losses increase, distortion in the microwave generator due to reflected power from load and damage to the transmitter can occur. 
         [0011]    The voltage standing wave ratio (VSWR) is a measure of how well a load is impedance-matched to a source. The value of VSWR is always expressed as a ratio with 1 in the denominator (2:1, 3:1, 10:1, etc.). It is a scalar measurement only (no angle), so although they reflect waves oppositely, a short circuit and an open circuit have the same VSWR value (infinity:1). A perfect impedance match corresponds to a VSWR of 1 (1:1). 
         [0012]    In regular antennas used during microwave ablation procedures there is not a maximum transfer of energy. For example, the regular dipole antenna without an integrated matching network can be modeled using the transmission line structure shown in  FIG. 4 . As shown in  FIG. 4 , the antenna  420  is coupled to a generator  400  via a coax cable  410  that has an impedance (Z c ) of 50Ω. In the presence of a half wave (λ/2) dipole antenna, the impedance (Z A ) of the antenna  420  is 73Ω according to dipole antenna theory. This results in a VSWR of 1.46 and, as such, some of the energy is not transferred to the antenna. 
       SUMMARY 
       [0013]    The present disclosure provides a microwave ablation system. The microwave ablation system includes a generator operable to output energy, an antenna coupled to the generator via a coaxial cable. The antenna is operable to deliver energy to tissue and includes a proximal radiating section and a distal radiating section. The antenna also includes a feedpoint section defined by the portion of the distal radiating section that underlaps the proximal radiating section. 
         [0014]    In another embodiment of the present disclosure, an antenna for use in a microwave ablation system is provided. The antenna is operable to deliver energy to tissue and includes a proximal radiating section and a distal radiating section. The antenna also includes a feedpoint section defined by the portion of the distal radiating section that underlaps the proximal radiating section. 
         [0015]    In other embodiments, the proximal radiating section is a semi rigid coaxial cable and/or the distal radiating section is a hypotube. 
         [0016]    In yet another embodiment, the distal radiating section is filled with a dielectric material having a first permittivity constant. The feedpoint section may also be filled with a dielectric material having the first permittivity constant. 
         [0017]    In other embodiments, the feedpoint section is filled with a dielectric material having a second permittivity constant different from the first permittivity constant. 
         [0018]    In other embodiments, an impedance of the feedpoint section is based on the first permittivity constant, an inner diameter of the proximal radiating section and the outer diameter of the distal radiating section. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0019]    The above and other aspects, features, and advantages of the present disclosure will become more apparent in light of the following detailed description when taken in conjunction with the accompanying drawings in which: 
           [0020]      FIG. 1  shows a representative diagram of a variation of a microwave antenna assembly in accordance with an embodiment of the present disclosure; 
           [0021]      FIG. 2  shows a cross-sectional view of a radiating portion of a microwave antenna assembly in accordance with an embodiment of the present disclosure; 
           [0022]      FIG. 3  shows a transmission line structure used to model a radiating portion of a microwave antenna assembly in accordance with an embodiment of the present disclosure; and 
           [0023]      FIG. 4  shows a transmission line structure used model a radiating portion of a dipole antenna without an integrated matching network. 
       
    
    
     DETAILED DESCRIPTION 
       [0024]    Particular embodiments of the present disclosure are described hereinbelow with reference to the accompanying drawings; however, it is to be understood that the disclosed embodiments are merely examples of the disclosure and may be embodied in various forms. Well-known functions or constructions are not described in detail to avoid obscuring the present disclosure in unnecessary detail. Therefore, specific structural and functional details disclosed herein are not to be interpreted as limiting, but merely as a basis for the claims and as a representative basis for teaching one skilled in the art to variously employ the present disclosure in virtually any appropriately detailed structure. Like reference numerals may refer to similar or identical elements throughout the description of the Fig.s. As used herein, the term “distal” refers to that portion of the instrument, or component thereof which is closer to the patient while the term “proximal” refers to that portion of the instrument or component thereof which is further from the patient. 
         [0025]    Electromagnetic energy is generally classified by increasing energy or decreasing wavelength into radio waves, microwaves, infrared, visible light, ultraviolet, X-rays and gamma-rays. As used herein, the term “microwave” generally refers to electromagnetic waves in the frequency range of 300 megahertz (MHz) (3×10 8  cycles/second) to 300 gigahertz (GHz) (3×10 11  cycles/second). As used herein, the term “RF” generally refers to electromagnetic waves having a lower frequency than microwaves. The phrase “ablation procedure” generally refers to any ablation procedure, such as RF or microwave ablation or microwave ablation assisted resection. The phrase “transmission line” generally refers to any transmission medium that can be used for the propagation of signals from one point to another. 
         [0026]      FIG. 1  shows a microwave antenna assembly  100  in accordance with one embodiment of the present disclosure. Antenna assembly  100  includes a radiating portion  12  that is connected by feed line  110  (or shaft) via cable  15  to connector  16 , which may further connect the assembly  10  to a power generating source  28 , e.g., a microwave or RF electrosurgical generator. Assembly  100 , as shown, is a dipole microwave antenna assembly, but other antenna assemblies, e.g., monopole antenna assemblies, may also utilize the principles set forth herein. Distal radiating section  105  of radiating portion  12  includes a tapered end  120  which terminates at a tip  123  to allow for insertion into tissue with minimal resistance. It is to be understood, however, that tapered end  120  may include other shapes, such as without limitation, a tip  123  that is rounded, flat, square, hexagonal, cylindroconical or any other polygonal shape. 
         [0027]    An outer jacket  124  is disposed about the outer cylindrical surface of antenna assembly  100 , e.g., the distal radiating section  105  and proximal radiating section  140 . Outer jacket  124  may be formed from any suitable material, including without limitation polymeric or ceramic materials. In some embodiments, outer jacket  124  is formed from PTFE. Outer jacket  124  may be applied to antenna assembly  100  by any suitable manner, including without limitation, heat shrinking. 
         [0028]      FIG. 2  shows a cross sectional view of radiating portion  12  of the antenna assembly  100 . In an embodiment of the present disclosure, a matching network is integrated into the radiating portion  12  at the feedpoint section  202  by underlapping the distal radiating section  105  into proximal radiating section  140  that extends distally from the feed line  110 . The inner conductor  204  is extended to distal end of the distal radiating section  105  and electrically coupled to the distal radiating section  105  thereby creating a short circuit. 
         [0029]    As shown in  FIG. 2 , the proximal radiating section  140  is a semi rigid coaxial cable with an outer conductor  206  and an inner conductor  204  extending through the proximal radiating section  140  into the distal radiating section  105 . Proximal radiating section is filled with a dielectric material  205  having a permittivity ∈ C  along the length L PRS  that results in a characteristic impedance (Z C ) for the proximal radiating section of 50Ω. The inner portion of the outer conductor  206  of the proximal radiating section  140  has a diameter D 1  as shown in  FIG. 2 . The inner conductor extending through the proximal radiating section  140  has a diameter d 2 . 
         [0030]    The distal radiating section  105  includes a TW hypotube  208  having a length L 2  with an outer diameter d 1  and an inner diameter of D 2 . The length L DRS  of the distal radiating section  105  extends from end  210  of the underlapping of the distal radiating section  105  and proximal radiating section  140  to tip  123 . Hypotube  208  is filled with a dielectric material  222  having a permittivity ∈ 2 . 
         [0031]    Feedpoint section  202  includes the underlapping of the distal radiating section  105  and the proximal radiating section  140 . The underlapping has a length L 1  which can be varied to optimize the feedpoint for maximum transfer of energy. Along length L 1 , the feedpoint is filled with a dielectric material  224  having a permittivity ∈ 1 . Dielectric materials  222  and  224  may have the same permittivity constant or different permittivity constants. 
         [0032]    The underlap forms a first coaxial line of length L 1  with a characteristic impedance of Z O1  which depends on the radial dimensions d 1  and D 1  and dielectric ∈ 1 . Z O1  can be calculated using the equation: 
         [0000]    
       
         
           
             
               Z 
               
                 O 
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                  
                 1 
               
             
             = 
             
               
                 138 
                 
                   
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                     1 
                   
                 
               
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                   Log 
                   10 
                 
                  
                 
                   ( 
                   
                     
                       D 
                       1 
                     
                     
                       d 
                       1 
                     
                   
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         [0033]    The wavelength (λ1) inside the first coaxial line can be calculated using the equation: 
         [0000]    
       
         
           
             
               
                 λ 
                 1 
               
               = 
               
                 c 
                 
                   f 
                    
                   
                     
                       ɛ 
                       1 
                     
                   
                 
               
             
             , 
           
         
       
     
         [0034]    where f equals frequency and c equals the speed of light. 
         [0035]    The inner conductor  204  and the distal radiating section  105  forms a second coaxial line terminating in a short circuit having a length L 2  and a characteristic impedance Z O2 . Z O2  can be calculated using the equation: 
         [0000]    
       
         
           
             
               Z 
               
                 O 
                  
                 
                     
                 
                  
                 2 
               
             
             = 
             
               
                 138 
                 
                   
                     ɛ 
                     2 
                   
                 
               
                
               
                 
                   Log 
                   10 
                 
                  
                 
                   ( 
                   
                     
                       D 
                       2 
                     
                     
                       d 
                       2 
                     
                   
                   ) 
                 
               
             
           
         
       
     
         [0036]    The wavelength (λ2) inside the first coaxial line can be calculated using the equation: 
         [0000]    
       
         
           
             
               
                 λ 
                 2 
               
               = 
               
                 c 
                 
                   f 
                    
                   
                     
                       ɛ 
                       2 
                     
                   
                 
               
             
             , 
           
         
       
     
         [0037]    where f equals frequency and c equals the speed of light, 
         [0038]      FIG. 3  shows a transmission line structure  310  that can be used to model the radiating portion  12  according to an embodiment of the present disclosure. In the model, the 50Ω coaxial cable or feed line is in series with a shorted stub  312  having a length L 2  and an impedance Z O2  and a piece of line  314  having a length L 1  and an impedance Z O1  all terminating on an antenna load  316  having an impedance Z A . In the matching network, the sum of Z 1  and Z 2  should equal the 50Ω coaxial cable in order to obtain a VSWR equal to 1 for maximum energy transfer. This can be accomplished by varying the lengths L 1  and L 2  and the dielectric permittivity constants ∈ 1  and ∈ 2  so that the sum of Z 1  and Z 2  equals or is close to the 50Ω coaxial cable. 
         [0039]    According to basic transmission line theory, L 1  can be selected so that Z 1  will have a resistance of 50Ω and a reactance of X 1 . Therefore, Z 1  would equal 50 Ω+jX 1 . L 2  can also be selected so that Z 2 =−jX 1 . Therefore, the sum of Z 1  and Z 2  would equal 50Ω. In an example according to an embodiment of the present disclosure, a 0.047″ coaxial cable can be used for proximal radiating section  140  having an inner conductor diameter (d2) of 0.011″ and an outer conductor inner diameter (D 1 ) of 0.037″. A 22 gauge TW hypotube can be used for the distal radiating section  105  having an inner diameter (D 2 ) of 0.020″ and outer diameter (d 1 ) of 0.028″. Teflon® can be used as dielectric material  224  having a permittivity (∈ 1 ) of 2.1. Accordingly, Z O1  would equal 11.5Ω. If L 1  was moved 0.018λ towards the generator, than the new normalized impedance (Z N1 ) would be equal to 4.35−j2.9 and Z 1 =Z N1 *Z O1 =50Ω−j33.5. At 915 MHz, λ 1 =0.226 m and L 1 =0.018λ 1 =4 mm. 
         [0040]    If Teflon® was also used as dielectric material  222 , impedance (Z O2 ) would be 24.7Ω. In order to obtain Z 2 =j33.5Ω, Z N2  would be 1.36Ω and L 2  would be 0.148λ which equals 3.3 cm. Since the antenna is a λ/2 dipole antenna that is typically 3.8 cm long at 915 MHz, the L DRS =1.9 cm. The distal radiating section  105  has a tapered end  120  with a sharp tip  123  that is typically 4-5 mm long. Assuming a length of the tip as 4 mm and a length of 4 mm for L 1 , the maximum length for L 2  is limited to 1.9 cm. Accordingly, the length of 3.3 cm needed to provide a perfect match with Teflon® is too long. If L 2  was 1.9 cm, L 2  would be 0.84λ resulting in a reactance of 0.59 which would results in Z 2  being j14.6Ω. As such, with Teflon® being used as dielectric material  222  and  224 , the sum of Z 1  and Z 2  would be 50Ω−j18.9 resulting in a VSWR of 1.46 which would be the same as an antenna without a matching network. 
         [0041]    However, if alumina was used for dielectric material  222 , (∈ 2 =9), λ2 would be 0.1093 m and Z O2  would equal 11.94Ω. In order for Z N2  to equal 2.81, L 2  would have to equal 2.1 cm which would still be too long as described above. Accordingly, if L 2  remains at 1.9 cm which equals 0.174λ, then the reactance would equal 1.93 and Z 2  would equal j23. The sum of Z 1  and Z 2  would be 50Ω−j10.5 which would result in a VSWR of 1.23. Therefore, the use of a combination of Teflon® and alumina would result in a better impedance matching and more energy would be transferred than an antenna without a matching network. 
         [0042]    It should be noted that different materials could be used than the ones listed in the examples above. By varying the dielectric materials used, the types of coaxial cable and hypotubes and the lengths L 1  and L 2 , a better impedance match may be obtained. Other dielectric materials that may be used include, but are not limited to, titanium dioxide, which has a permittivity ranging from 10-40, or other ceramic materials having a similar permittivity. Low density Teflon® may also be used instead of regular Teflon®. 
         [0043]    The described embodiments of the present disclosure are intended to be illustrative rather than restrictive, and are not intended to represent every embodiment of the present disclosure. Various modifications and variations can be made without departing from the spirit or scope of the disclosure as set forth in the following claims both literally and in equivalents recognized in law.