Abstract:
A method of processing a signal in a wireless digital communications system, wherein a source of disturbance affects differently at least first and second portions of a received signal carrying user data and/or control data, the method comprising: identifying the second portion of the received signal, most affected by the source of disturbance; generating a first estimate of the disturbance (P I ) for received samples in the first, less affected portion of the received signal; generating a second estimate of the disturbance (P I(SCH) ) for received samples in the second portion of the received signal; and using the first and second disturbance estimates to generate reliability information for the data bits corresponding to the received signal samples, for use in a decoding process to estimate the transmitted data bits.

Description:
FIELD OF THE INVENTION 
     This invention relates to processing in a wireless digital communication system and in particular where there is a source of non-stationary interference. 
     BACKGROUND OF THE INVENTION 
     In a 3GPP WCDMA (Wide Band Code-Division Multiple Access) digital communication system a Synchronisation Channel (SCH) is transmitted by base station equipment (Node B) of a cell for the first 256 chips of each 2560 chip time slot (3GPP TS 25.211, “Technical Specification Group Radio Access Network: Physical Channels and Mapping of Transport Channels onto Physical Channels (FDD)”, June 2005, Section 5.3.3.5). The data transmitted on the SCH is essential for a User Equipment (UE) receiver in order to synchronise to the cell. As it is such an important channel, the SCH may be transmitted at a high power relative to other downlink channels transmitted in the cell. Unlike other WCDMA downlink channels, the SCH is not encoded using an Orthogonal Variable Spreading Factor (OVSF) code and therefore, even in the absence of multipath fading conditions, it is not orthogonal to other channels in the cell. When the user equipment is operating in high cell geometry (i.e. it is close to the centre of the cell), the SCH may be a significant source of interference. This may be problematic for two reasons. Firstly, the SCH signal may affect the received signals of other lower power downlink channels, causing errors of phase and amplitude in the corresponding data samples. Secondly, interference estimates made on specific chip positions within the slot may not correctly take the SCH interference into account. For downlink Dedicated Physical Channels (DPCHs), the interference experienced during a slot is typically estimated based on the DPCH dedicated pilot symbols transmitted at the end of the slot, or based on the Common Pilot Channel (CPICH) transmitted over the entire slot. When the interference is estimated from the dedicated pilots, depending on the relative delay between the DPCH and SCH, the SCH may affect the DPCH data fields but not the pilot field. When the interference is estimated from the CPICH, the SCH only affects a single CPICH symbol of the CPICH slot. Therefore, neither method can correctly take into account the SCH interference on the affected data samples. As a result, the DPCH symbols affected by the SCH will have a much higher actual level of interference than that estimated. 
       FIG. 1  is a schematic block diagram of the receiver for a digital communications system. An antenna  2  receives an incoming signal  4  over a wireless communication channel. The incoming signal  4  is supplied to RF and IF stages  6  which provide a baseband signal to a receiver front end  8  where analogue to digital conversion takes place. 
     The receiver front end  8  supplies digitised data to a signal detector  10  which generates received signal samples y k  for further processing in the receiver. In a wide band code-division multiple access user equipment, the signal detector  10  can take the form of a rake receiver or a chip level equaliser with suitable descrambling and despreading components. These generate DPCH signal samples y k  in a known manner. 
     One of the functions of the receiver is to calculate reliability information on the received data bits, for example in the form of bit log likelihood ratios (LLRs). An LLR calculation block is denoted by reference numeral  12  in  FIG. 1 . The log likelihood ratios are supplied to a deinterlever and channel decoder  14  for decoding the received data bits associated with the signal samples y k . Channel decoders which take into account reliability information for example in the form of log likelihood ratios are often referred to as soft-input channel decoders. Examples are given by soft-input/hard-output convolutional decoders or soft-input/soft-output turbo decoders. 
     It can be seen that in situations where the SCH interference (which can be significant on the affected data samples) is not properly taken into account, not only the data bits corresponding to these signal samples will be subject to higher interference, but their reliability estimates will indicate a much higher level of reliability than is in fact the case. 
     The DPCH samples y k  are modelled as the sum of a signal component and an interference-plus-noise component
 
 y   k   =a √{square root over ( E   s )} s   k   +n   k ,  Equation 1
 
where s k =s Ik +js Qk , s Ik =b 1k , s Qk =b 2k  ε{+1,−1} are the QPSK symbols transmitted on the DPCH, b 1k , b 2k  denote the bits mapped onto each symbol, aεR +  wherein R +  is the set of positive real numbers, E s  is the received symbol energy, and n k =n Ik +jn Qk  represents the noise-plus-interference, modelled as an additive complex Gaussian process with zero mean and variance N 0 . The bit LLRs relative to the received signal r k  can be derived independently for each of the two bits b 1k , b 2k  mapped to the QPSK symbol s k . Considering the bit b ik , i=1, 2 and letting y ik =Re[y k ] for i=1 and y ik =Im[y k ] for i=2, we have
 
     
       
         
           
             
               
                 
                   
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       FIG. 2  is a schematic block diagram of the LLR calculation block  12 . As shown in  FIG. 2 , the LLR for the DPCH bits is calculated using an estimate of the received signal energy E s  and of the noise-plus-interference variance N 0  (block  16 ). Specifically, apart from a suitable scaling factor η, the LLR is obtained as the product of the demodulated signal amplitude y ik  and an estimate of the quantity √{square root over (E s )}/N 0  in the LLR scaling block  18   
     
       
         
           
             
               
                 
                   
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     As illustrated in  FIG. 2 , the estimate of the quantities E s  and N 0  is obtained from the dedicated DPCH pilots corresponding to the samples y k  at specified positions within the DPCH time slot (normally at the end). Since the DPCH time slot is transmitted with a specified delay τ DPCH  with respect to the SCH time slot, the DPCH dedicated pilot symbols may not be affected by the SCH. If this happens, and if the SCH represents a significant portion of the total received power, then the received DPCH symbols corresponding to the position of the SCH may have a very large amplitude due to SCH interference, and at the same time the estimate of the noise-plus-interference power N 0  may be small, because it does not take into account the SCH interference. In this situation, the reliability of the received data will be overestimated, potentially by a significant amount. Large, erroneous LLRs will have a negative impact on the channel decoder&#39;s ability to correct errors. 
     An alternative possibility is to base the estimate of N 0  on samples y k(CPICH)  of CPICH symbols distributed throughout the slot. In this case, E s  can be estimated using the DPCH pilot samples. If the estimate of N 0  is based on the CPICH symbols, and is obtained from the entire CPICH time slot, again for the received DPCH symbols corresponding to the position of the SCH the noise-plus-interference power will be underestimated, and the reliability of the received data will be overestimated. 
     The fact that the presence of the SCH channel may limit the performance of the WCDMA downlink has been already discussed in an article by F. Kaltenberger, K. Freudenthaler, S. Paul, J. Wehinger, C. F. Mecklenbräuker and S. Springer, “Throughput enhancement by cancellation of synchronisation and pilot channel for UMTD High Speed Downlink Packet Access”, Proceedings of 6 th  IEEE International Workshop on Advances in Wireless Communications (SPAWC), New York, USA, June 2005, pp. 603-607. This article proposes to resolve the problem by cancelling the interference due to the SCH. This approach has the disadvantage of a high implementation cost due to the complexity of the required circuitry in the receiver. 
     SUMMARY OF THE INVENTION 
     According to an aspect of the present invention there is provided a method of processing a signal in a wireless digital communications system, wherein a source of disturbance affects differently at least first and second portions of a received signal carrying user data and/or control data, the method comprising: identifying the second portion of the received signal most affected by the source of disturbance; generating a first estimate of the disturbance (P I ) for received samples in the first, less affected portion of the received signal; generating a second estimate of the disturbance (P I(SCH) ) for received samples in the second portion of the received signal; and using the first and second disturbance estimates to generate reliability information for data bits corresponding to the received signal samples, for use in a decoding process to estimate the transmitted data bits. 
     Another aspect of the invention provides a receiver for processing a signal in a wireless digital communications system, wherein a source of disturbance affects differently at least first and second portions of a received signal carrying user data and/or control data, the receiver comprising: a component adapted to identify the second portion of the received signal, most affected by the source of disturbance; a first estimator adapted to generate a first disturbance estimate (P I ) for received samples in the first, less affected portion of the received signal; a second estimator adapted to generate a second disturbance estimate (P I(SCH) ) for received samples in the second, most affected portion of the received signal; and a reliability information generator adapted to use the first and second disturbance estimates to generate reliability information for data bits corresponding to the received signal samples, for use in a decoder to estimate the transmitted data bits. 
     A further aspect provides a digital communications system comprising: an antenna for receiving a wireless signal; a receiver front-end for receiving the wireless signal in analogue form and providing digital samples therefrom; a baseband receiver for processing the signal samples, said baseband receiver comprising: a component adapted to identify a second portion of the received signal, most affected by a source of disturbance; a first estimator adapted to generate a first disturbance estimate for received samples in a first, less affected portion of the received signal; a second estimator adapted to generate a second disturbance estimate for received samples in the second portion of the received signal; a reliability information generator adapted to use the first and second disturbance estimates to generate reliability information for data bits corresponding to the received signal samples; and a decoder for receiving the reliability information and arranged to estimate the transmitted data bits. 
     A further aspect provides an integrated circuit incorporating a receiver for processing a signal in a wireless digital communications system, wherein a source of disturbance affects differently first and second portions of a received signal carrying user data and/or control data, the receiver comprising: a component adapted to identify the second portion of the received signal most affected by the source of disturbance; a first estimator adapted to generate a first disturbance estimate for received samples in the first, less affected portion of the received signal; a second estimator adapted to generate a second disturbance estimate for received samples in the second portion of the received signal; and a reliability information generator adapted to use the first and second disturbance estimates to generate reliability information for data bits corresponding to the received signal samples, for use in a decoder to estimate the transmitted data bits. 
     A further aspect provides a mobile terminal incorporating a receiver for processing a signal in a wireless digital communications system as defined in the foregoing. 
     A further aspect provides a computer program product comprising computer code means which when installed in a computer implements the following steps in a method of processing a signal in a wireless digital communications system, the method steps comprising: identifying a second portion of the received signal most affected by the source of disturbance; generating a first estimate of disturbance for received samples in a first less affected portion of the received signal; generating a second estimate of the disturbance for received samples in the second portion of the received signal; and using the first and second disturbance estimates to generate reliability information for data bits corresponding to the received signal samples, for use in a decoding process to estimate the transmitted data bits. 
     It will be appreciated that the word “disturbance” used herein refers to any part of the signal which is not information. That is, the disturbance could be due to noise or interference or a combination of the two. 
     The following described embodiments of the invention address the problem of computing the reliability information of data affected by interference that is present only over a limited time interval. The example which is discussed in the described embodiments is the case of a 3GPP WCDMA system, where the interference associated with downlink synchronisation SCH channels only affects a fraction of the data transmitted in a time slot, referred to as an affected portion. The remainder of the data is referred to as the unaffected portion. The method described herein makes use of, in one embodiment, a modified algorithm for the estimation of the total interference-plus-noise power, which relies on the identification of time intervals over which the interference can be modelled as a stationary noise process. 
     Although the specific example which is discussed in the following relates to this context, the invention is more broadly applicable. For example, techniques described herein can be used in other situations where “non-stationary” interference is present in a wireless digital communications system. Non-stationary interference is interference whose statistics (e.g. power) are not constant in time. However, there are situations where a non-stationary interference process can be modelled as practically stationary over a given time interval—with different statistics over different time intervals. For example, the techniques described herein could be used in a CDMA system where some of the channels (that are not synchronous with the channel containing the dedicated/common pilots) are not always transmitted and/or are transmitted with different power levels over different transmission intervals. 
     The following described embodiments of the invention provide a significant advantage with respect to the prior art discussed above, avoiding the need to cancel the interference due to the synchronisation channels. Instead, this interference is taken into account to properly compute the bit reliability for the affected data symbols. In addition to reducing the implementation cost, the solution of the present invention is computationally more robust. 
     For a better understanding of the present invention and to show how the same may be carried into effect, reference will now be made by way of example to  FIGS. 3 to 7  of the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a schematic block diagram of data flow in a receiver; 
         FIG. 2  is a schematic block diagram of calculation of reliability information for channel decoding; 
         FIG. 3  is a schematic block diagram of one embodiment of the invention; 
         FIG. 4  is a schematic diagram illustrating a channel affected by interference from a synchronisation channel; 
         FIG. 5  is a schematic block diagram of a second embodiment of the invention; 
         FIG. 6  is a schematic block diagram of a third embodiment of the invention; 
         FIG. 7  is a schematic diagram illustrating a channel affected by interference from a plurality of synchronisation channels; and 
         FIGS. 8A-8D  illustrate four different sets of symbols affected in different ways by three SCH signals. 
     
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       FIG. 3  is a schematic block diagram of a first embodiment of the present invention arranged to address the problem of degraded channel decoding performance in the presence of high SCH interference in a 3GPP WCDMA digital communications system. Before discussing operation of  FIG. 3 , reference will first be made to  FIG. 4  which schematically illustrates the relationship between SCH and DPCH channels in a WCDMA digital communications system. 
       FIG. 4  shows a transmission slot in a WCDMA digital communications system. In the described embodiment, the slot contains 2560 chips. There are 15 slots per 10 ms transmission frame. On the SCH channel, transmission takes place only for the first 256 chips of each slot, which carry the synchronisation signal referenced  20  in  FIG. 4 . 
     As is know in the art, the DPCH slot can carry user data from one or more transport channels in two different data fields, Ndata 1  and Ndata 2 . The time slot in the DPCH channel also carries pilot symbols  22  at the end of the slot. The portion of the DPCH slot in  FIG. 4  carrying user data is referred to as Dedicated Physical Data Channel (DPDCH). The present invention also applies to the portion of the DPCH slot in  FIG. 4  carrying control data, referred to as Dedicated Physical Control Channel (DPCCH), including power control command (TPC) symbols. It can be seen from  FIG. 4  that the DPCH symbols affected by the SCH depend on the delay τ DPCH  which is defined as the DPCH timing offset with respect to the P-CCPCH frame. The delay τ DPCH  effectively coincides with the timing offset with respect to the SCH frame, and so allows the location of symbols affected by the SCH signal to be determined. The slot format of the DPCH is defined in 3GPP TS 25.211, “Technical Specification Group Radio Access Network; Physical Channels and Mapping of Transport Channels onto Physically Channels (FDD)”, June 2005, Section 5.3.2. 
     As shown in  FIG. 4 , depending on the value of the timing offset τ DPCH , the bits affected by SCH interference may be in the DPDCH fields Ndata 1  or Ndata 2 . Although not shown in  FIG. 4 , the SCH interference may also affect the DPCCH fields. In the example of  FIG. 4 , in neither case are the pilot symbols affected. It is noted that in the event that only the dedicated pilot symbols are affected by SCH interference, there is no need to use the techniques described herein. In fact, in that case, the disturbance estimated on the pilot symbols will be higher than that experienced by the rest of the slot, resulting in LLRs which are uniformly depressed from their true value (that is, the one they would have if disturbance estimation were perfect) The uniformity of this change means that the decoder performance is not severely affected and so no special action will be required. 
     Reference will now be made to  FIG. 3  to describe one embodiment of the invention.  FIG. 3  illustrates in schematic block diagram form circuitry at a receiver in a wireless digital communication system. The signal detector  10  takes the same form as that shown in  FIG. 1 . This is known in the art and so will not be described further. As is also known, a block  16  which estimates received signal energy E s  and noise-plus-interference variance N 0  is arranged to receive dedicated DPCH pilot samples  22  from the incoming samples y k . 
     An affected sample determination block  24  determines symbols which are affected by the SCH signal  20  using the delay τ DPCH . The set of affected samples is denoted {y k } SCH . The affected samples are supplied to a power estimation block  26 . The power estimation block  26  estimates the average total received power P T  from the set of samples {y k } SCH  corresponding to the N SCH  DPCH symbols affected by the SCH signal  20 : 
     
       
         
           
             
               
                 
                   
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     Using a signal power estimate P S  derived from the DPCH dedicated pilot symbols (from the symbol energy estimate E s ), the power estimation block determines the interference power P I(SCH)  on the received symbols affected by the SCH according to
 
 P   I(SCH)   =P   T   −P   S .  Equation 5
 
     A maximum power value P′ I(SCH)  is computed in block  28  as the maximum between P I(SCH)  and the interference power P I  for the received symbols not affected by the SCH (originally calculated as N 0  by using the DPCH dedicated pilot symbols):
 
 P′   I(SCH) =max( P   I(SCH)   ,P   I )  Equation 6
 
     For the QPSK modulated DPCH, LLR values L(b ik |y k ) are then calculated by a first LLR scaling function FN 1  in scaling block  30  using N 0 =P′ I(SCH)  for the symbols affected by the SCH: 
     
       
         
           
             
               
                 
                   
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     A second LLR scaling function FN 2  in block  30  calculates LLR values using N 0 =P I  for the symbols not affected by the SCH: 
     
       
         
           
             
               
                 
                   
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     The LLRs are then processed by deinterleaving, depuncturing and soft-input channel decoding (block  34 ), which uses the reliability information on the coded bits to derive an estimate of the transmitted uncoded bits in a manner which is known in the art and so will not be discussed further herein. 
       FIG. 5  illustrates an alternative embodiment of the invention which avoids the need for two separate LLR scaling functions as in  FIG. 3 . In  FIG. 5  like numerals denote like parts as in  FIG. 3  and perform the same function. These parts will not be described further. A ratio determining block  38  determines a ratio of P I  to P′ I(SCH)  and this ratio is supplied to a multiplier  40  which multiplies affected samples {y k } SCH  by the ratio. The multiplied samples are then supplied to an LLR scaling block  18  which can then operate similarly as in  FIG. 2 , by performing the LLR calculation using N 0 =P I . The effect of the multiplication is that the outcome of the LLR calculation using N 0 =P I  is equivalent to that produced by equation 7 and equation 8: 
     
       
         
           
             
               
                 
                   
                     
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       FIG. 6  illustrates an alternative embodiment where the estimation of the noise-plus-interference power is based on the CPICH common pilot symbols. As shown in  FIG. 6 , a select CPICH symbol block  42  identifies which CPICH symbols are affected by the SCH signal  20  and divides the CPICH symbols into two subsets, a subset  44  which is affected and a subset  46  which is not affected. Each subset is supplied to a respective N 0  estimation block  48 ,  50  which derive estimates for the noise-plus-interference power N 0  for each set. These are denoted N′ 0(SCH)  for the affected set and N′ 0  for the non-affected set. The estimate of the noise-plus-interference power N′ 0(SCH)  for the CPICH symbols affected by the SCH signal  20  is applied from the block  48  to an LLR scaling function FN 2 ′. Similarly, the N′ 0 , values for the non-affected symbols are supplied to an LLR scaling function FN 1 ′ for the non-affected symbols. The average signal energy E s  is determined from DPCH pilot samples y k  as from  FIGS. 3 and 5 . 
       FIG. 7  illustrates a version of the embodiment of  FIG. 6  but which requires only one LLR scaling function (as in  FIG. 5 ). Like numerals denote like parts as in  FIG. 6  and its operation will readily be appreciated from the description of  FIGS. 5 and 6 . 
     Note that in addition to duplicating the elements  26 ,  28 ,  38  it is necessary to estimate an average noise plus interference variance N 0 —see block  19 . 
     The usefulness of the above techniques can be intuitively understood in the following terms. In the case where the SCH interference is low compared with other sources of interference, then this procedure has no practical effect on the LLR values for those symbols which coincide with the SCH. However, where the SCH interference is large, using a value for the noise-plus-interference variance which takes this into account (e.g. N 0 =P′ I(SCH)  in  FIG. 3 ) in the LLR calculation ensures that the bits corresponding to symbols affected by the SCH have smaller LLRs than their unaffected counterparts, which means in turn that they will carry less weight during the decoding process being correctly identified as less reliable. 
     A suboptimal, but lower complexity, implementation may be achieved by introducing erasures in the positions of the strongly affected symbols. This approach may be practically adequate for high levels of SCH interference, but the full algorithm allows for a precise calculation of the bit reliability for intermediate levels of SCH interference. 
     For a UE in soft handover or softer handover, i.e. in the case where the UE is receiving the same information in the downlink from multiple cells, (see 3GPP TS 25.922, “Technical Specification Group Radio Access Network; Radio Resource Management Strategies”, December 2006), there is the possibility that the received DPDCH signal is affected by more than one downlink SCH transmission.  FIG. 8  illustrates four sets of symbols affected in different ways by three SCH signals.  FIGS. 8   a  through  8   c  illustrate three SCH channels with the SCH signals illustrated with respect to a time t=0 which denotes the beginning of a slot for the SCH channel  1 . 
       FIG. 8   d  illustrates a DPCH slot showing two data fields Ndata 1 , Ndata 2 . The result is four sets of symbols affected in different ways by the three SCH signals. 
     That is, a first set of symbols in Ndata 1  is affected only by SCH channel  1 , a second set of symbols in Ndata 1  is affected by SCH channel  1  and SCH channel  2 , and a third set of symbols in Ndata 2  are affected by SCH channel  3 . Which symbols are affected in this case can be determined using the values of τ DPCH  for each cell. The algorithm may be adapted either by calculating a value for P′ I(SCH)  for each set of symbols affected by the same set of SCHs (see  FIG. 3 ), or by only considering the SCH from the strongest cell (considering that it is unlikely that a UE will be sufficiently close to multiple cells for more than one SCH to interfere significantly). 
     Detailed computer simulations have been carried out to assess the effect of the proposed method for noise-plus-interference power estimates obtained from the DPCH dedicated pilots. The assessment has been based on a measure of the DPCH block error rate (BLER) under AWGN propagation conditions, and a single downlink cell with cell geometry of 35 dB (cell geometry being defined as the ratio between the total downlink power received from the wanted cell and the total power received from other cells plus thermal noise). The simulations refer to the transmission of two multiplexed DCH Transport Channels (TrCHs) in fixed positions, with the following parameters (3GPP TS 25.212, “Technical Specification Group Radio Access Network; Multiplexing and Channel Coding (FDD)”, June 2005):
         TrCH  1 : channel coding=turbo code, transport block size=672 bits, transmission time interval=20 ms, rate matching attribute=135.   TrCH  2 : channel coding=convolutional code, transport block size=148 bits, transmission time interval=40 ms, rate matching attribute=185.       

     The downlink DPCH has been transmitted using slot format  12  (3GPP TS 25.211, “Technical Specification Group Radio Access Network; Physical Channels and Mapping of Transport Channels onto Physical Channels (FDD)”, June 2005, Section 5.3.2), with transmit power determined by the WCDMA power control algorithm (3GPP TS 25.214, “Technical Specification Group Radio Access Network; Physical Layer Procedures (FDD)”, June 2005). The sum of the Primary SCH (P-SCH) and Secondary SCH (S-SCH) transmit power has been set 6 dB higher than the CPICH power. No Orthogonal Channel Noise Simulator (OCNS) has been included in the downlink signal (3GPP TS 25.101, “Technical Specification Group Radio Access Network; User Equipment (UE) Radio Transmission and Reception (FDD)”, September 2005). 
     For τ DPCH =0 chips, the SCH position corresponds to the first 8 bits of every slot. The manner in which the channels TrCH 1 , TrCH 2  map onto the data fields Ndata 1 , Ndata 2  in a DPCH slot means that it affects only the turbo coded TrCH. With a conventional noise-plus-interference estimation, the turbo coded TrCH has experienced a BLER=15.6% and the convolutionally coded TrCH a BLER=0%. Using the proposed algorithm described with reference to  FIG. 5  (SCH samples are pre-scaled by P I /P I(SCH)  before being processed into LLRs using √{square root over (E s )}/P I ), the turbo coded TrCH performance has been seen to dramatically improve to a BLER=0%, with no effect on the BLER performance of the convolutionally encoded TrCH. 
     For τ DPCH =512 chips, the SCH position corresponds to the last 8 DPDCH bits of every slot, and the mapping of the channels to the slot means it affects both the turbo coded TrCH and the convolutionally coded TrCH. Without the proposed algorithm the turbo coded TrCH has experienced a BLER=5.4% and the convolutionally coded TrCH a BLER=22.6%. Using the proposed algorithm, both the turbo coded and convolutionally coded BLERs have been seen to reduce to 0%. 
     While the invention has been described in the context of the above-referenced embodiments, it will be appreciated that alternatives are possible, and that the scope of this invention is limited only by the accompanying claims.