Abstract:
A method to discriminate a real echo peak from an aliased echo peak comprises: computing ‘a correlation between N samples of a digital data and a copy of the same N samples delayed by a time delay Δ 1,0  to obtain a first correlation result, time delay Δ 1,0  being equal to a time interval T between a first and a second distinct power peaks of an estimated channel impulse response, the first peak being the highest power peak within the temporal window, and/or computing—a correlation between the N samples and a copy of the same N samples delayed by a time delay T IFFT -Δ 1,0  to obtain a second correlation result, and deciding whether the second peak is a real echo peak or an aliased echo peak based on the first and/or second correlation results.

Description:
FIELD OF THE INVENTION 
     The present invention relates to a method, a discriminator and a receiver to discriminate real echo peaks from aliased echo peak, OFDM symbol synchronization and equalizer tuning method using this method. 
     BACKGROUND OF THE INVENTION 
     There exist methods to cancel aliased echo peaks created by aliasing phenomenon according to Shannon theory in an estimated channel impulse response of a telecommunication channel. 
     For example, US 2005/0213680 in the name of Atungsiri et al. describes such a method to cancel aliased echo peak in an estimated channel impulse response of a channel used in an OFDM (Orthogonal Frequency Division Multiplexing) telecommunication system. 
     One can also refer to this document to obtain more information on OFDM receivers. 
     In wireless communication system, the wireless signal emitted by an emitter may follow different paths before reaching an antenna of a receiver. In this specification, the path through which is received the signal with the highest power is called the main path. The other paths are called secondary paths. 
     The signal that follows a secondary path is called an echo. 
     Typically, the main path and the secondary paths do not have the same length. 
     Thus, the signal is received at time t 1  when it follows the main path and at time t 2  when it follows the secondary path. Time t 2  may be in advance on time t 1  and in this case the echo is called a “pre-echo” or behind time t 1  and in this case the echo is called a “post-echo”. 
     In a real channel impulse response (CIR), an echo corresponds to an echo power peak at a time distinct from the time where appears the highest power peak corresponding to the signal that follows the main path. This echo power peak is called an echo peak in this specification whereas the highest power peak corresponding to the main path is called the main peak. 
     An echo peak is strongly correlated to the main peak because the echo peak is created by a copy of the signal that creates the main peak, but with an offset in time. 
     As explained in US 2005/0213680, aliased echo peaks can appear in an estimation of the channel impulse response (CIR) when the estimation is built from an insufficient number of sampled data. This happens according to the well-known Shannon&#39;s theory. The estimation of a channel impulse response is built over a specified temporal window. In OFDM communication systems, the temporal window is known as IFFT (Inverse Fast Fourier Transform)-window, for example. 
     Thus, it should be understood that an aliased echo peak corresponds to a real echo peak but is not placed at the right place in the temporal window over which the channel impulse response is estimated. From this aspect, an aliased echo peak is distinct from power peaks known as “ghost peak” or “replica peak” or “image peak” that do not correspond to a real echo. In fact, ghost peak, replica peak and image peak are generated by parasitic effects like Doppler effect but not by echo. This is an important difference because to correct the estimated channel impulse response, the ghost peak and the like have to be cancelled whereas the aliased echo peaks have to be replaced at the right place in the estimated channel impulse response. 
     However, the aliasing of an echo peak only appears if the time interval T between the echo and the signal that follows the main path is greater than a predetermined limit. In fact, if time interval T is inferior to this predetermined limit, the echo peak appears at the right position in the estimated channel impulse response. In this situation, the echo peak is called real echo peak in this specification. 
     In contrast, if time interval T is greater than the predetermined limit, aliasing occurs and the estimated channel impulse response includes an aliased echo peak instead of a real echo peak. 
     Most of the time, the value of time interval T is not known. Thus, it is difficult to discriminate aliased echo peaks from real echo peaks. 
     SUMMARY OF THE INVENTION 
     Accordingly, it is an object of the invention to provide a method to discriminate a real echo peak from an aliased echo peak in an estimated channel impulse response of a telecommunication channel. The invention is defined by the independent claims. Dependent claims describe advantageous implementations. 
     The invention provides such a method wherein the method comprises: 
     sampling a received analogic signal to obtain digital data corresponding to the received analogic signal, 
     estimating the channel impulse response over a temporal window of size T IFFT  from only a part of the digital data so that aliased echo peaks may appear in the estimated channel impulse response, 
     computing a correlation between N samples of the digital data and a copy of the same N samples delayed by a time delay Δ 1,0  to obtain a first correlation result, time delay Δ 1,0  being equal to a time interval T between a first and a second distinct power peaks of the estimated channel impulse response, the first peak being the highest power peak within the temporal window, and/or d) computing a correlation between the N samples and a copy of the same N samples delayed by a time delay T IFFT -Δ 1,0  to obtain a second correlation result, and 
     deciding whether the second peak is a real echo peak or an aliased echo peak based on the first and/or second correlation results. 
     The above method uses the fact that a time interval T 2  between an aliased echo peak and the main peak is different from a time interval T 3  between the main peak and the real echo peak corresponding to this aliased echo peak. Thus, the result of the correlation of the received signal with a copy of the same received signals but delayed by T 2  should be nearly null. In contrast, the result of a correlation of the received signal with a copy of the same received signal delayed by T 3 , should be far from zero. As a result, it is possible to discriminate a real echo peak from an aliased echo peak from those results of correlation. 
     The embodiment of the above terminal may comprise one or several of the following features: 
     the sampling frequency used in step a) corresponds to a sampling period T e , and step c) or d) is also iterated with other time delays equal to T±i·T e , where  i  is a non-zero positive integer and “·” is the symbol of the multiplication so as to obtain a plurality of first or second correlation results, and wherein during step e), it is decides whether the second peak is a real or an aliased echo peak based on the plurality of first or second correlation results, 
     the estimation of the channel impulse response is carried out from predetermined pilots present in OFDM (Orthogonal Frequency Division Multiplexing) symbols, the predetermined pilots being arranged with the OFDM symbols at frequency interval corresponding to  n  carrier frequencies and their position being shifted by  k  carrier frequencies from one OFDM symbol to the following one so that m*k=n, m, n and k being integer numbers greater than one, and wherein T IFFT  is equal to m/n*T u , where T u  is the duration of the modulation of an OFDM symbol, 
     the N samples are temporal successive digital data and N is chosen so that N·T e  is greater than or equal to k·T u , and preferably greater than or equal to 16·T u , and 
     the method comprises both steps c) and d), and wherein if the first correlation result or the plurality of first correlation results are greater than the second correlation result or the plurality of second correlation results, then it is decided that the second peak is a real peak, else it is decided that the second peak is an aliased peak. 
     The above embodiments of the terminal present the following advantages: 
     using a plurality of first and second correlation results increases the reliability of the method in particular when the received echo is highly spreaded 
     choosing N so that N·T e  is greater than or equal to 4 T u  in an OFDM system increases the reliability of the correlation result. 
     The invention also relates to a fine OFDM symbol synchronization method as well as an equalizer tuning method, which uses the above discriminating method. 
     The invention also relates to a discriminator and a receiver implementing the above method. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a schematic diagram of the structure of a terminal having an OFDM symbol receiver, 
         FIG. 2  is a flowchart of a fine OFDM symbol synchronization method, 
         FIG. 3  is a draft of an estimated channel impulse response, and 
         FIGS. 4A and 4B  are draft of a signal through a main path and of an echo of this signal. 
     
    
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS 
       FIG. 1  shows a DVB-T (Digital Terrestrial Video Broadcasting) terminal  2 . For example, terminal  2  is a mobile phone or a Set top Box. 
     Terminal  2  is adapted to receive wireless signals according to an OFDM communication protocol. The signal is a multi-carrier signal used to transport OFDM symbols. 
     The structure of terminals to receive OFDM symbols is well known so, for simplicity,  FIG. 1  shows only the details necessary to understand the invention. Terminal  2  has an antenna  4  to receive an analogic wireless signal and an OFDM symbol receiver  6  connected to antenna  4  through an input  8 . For example, receiver  6  is a radio frequency receiver which outputs a digital bit stream corresponding to the received signal through an output  10 . 
     Receiver  6  has an analogue-to-digital converter  12  connected to input  8  to obtain digital data corresponding to the received analogic wireless signal. Converter  12  samples the received analogic signal with a frequency f e  that corresponds to a sampling period T e . 
     The obtained digital data are sent to a Fast Fourier Transformer  14  and a coarse synchronizer  16 , inputs of which are connected to an output of converter  12 . 
     Transformer  14  is designed to perform FFT (Fast Fourier Transform) on the received signal during a FFT-window. 
     Synchronizer  16  is intended to perform a coarse OFDM symbol synchronization that consists in estimating the FFT-window position accurately enough so that post-FFT operations can be performed. 
     Synchronizer  16  outputs a coarse tuning instruction to a FFT-window positioning module  18 . Module  18  tunes the FFT-window position of transformer  14  according to the coarse FFT-window position estimation as well as, if available, to an estimated channel impulse response (CIR). More precisely, module  18  is able to fine-tune the position of the FFT-window according to the position of the peaks in the estimated channel impulse response. Then, module  18  outputs a tuning instruction to transformer  14  to fine-tune the FFT-window position. 
     Post-FFT operations relate to operations carried out on the symbols in the frequency domain outputted by transformer  14  like forward error correction, for example. 
     An equalizer  20  that equalizes the digital data in the frequency domain is connected at an output of transformer  14 . Equalizer  20  is a tunable equalizer having at least one coefficient that is automatically tuned according to an estimation of the channel impulse response of the channel through which is received the wireless signal. 
     Receiver  6  has a channel impulse response estimator  22  to build the estimated channel impulse response from scattered pilots present in the received signal. An input of estimator  22  is connected to an output of transformer  14  to receive OFDM symbols in the frequency domain. Estimator  22  is able to output the estimated channel impulse response to module  18  and equalizer  20 . 
     More precisely, pilots are predetermined symbols that are known by receiver  6  before they are received. Pilots are repeatedly sent in the signal so that the receiver can build an estimated channel impulse response. For example, scattered pilots are continuously sent during all the reception of these signals. For instance, in DVB-T standard, one predetermined pilot is shifted by k carrier frequencies from one symbol to the next one so that it is sent on the same frequency carrier every  m  OFDM symbols having pilots. Within one OFDM symbol, predetermined pilots are arranged on carrier frequencies, which are spaced apart by  n  carrier frequencies. Thus  m * k =n, where  m ,  k  and  n  are integer greater than one. For example,  m  equal 4, k equal 3, and  n  equal 12. This is a well-known process and will not be described in further details. 
     Receiver  6  has a discriminator  30  able to discriminate a real echo peak from an aliased echo peak in the channel impulse response outputted by estimator  22 . To do so, discriminator  30  has a plurality of autocorrelator  32   a  to  32   c  and  34   a  to  34   c . 
     Each of these correlators is able to correlate  N  samples of the digital data in the time domain, i.e. before transformer  14 , with a copy of the same  N  samples delayed by a predetermined time interval. To this end, discriminator  30  is connected to an input of transformer  14  and to estimator  22 . 
     The operation of receiver  6  will now be described with reference to  FIGS. 2 ,  3  and  4 . 
     Initially, in step  40 , the converter  12  samples the received analogic wireless signal and the converter  12  outputs corresponding digital data. 
     In step  44 , synchronizer  16  computes a coarse position for the FFT-window and outputs it to module  18 . Coarse synchronization may be done according to the method disclosed in the patent application WO 2005/002164. 
     Then, in step  46 , transformer  14  performs a Fast Fourier Transform of the received signal during the time interval defined by the FFT-window and outputs the received OFDM symbols in the frequency domain. 
     Subsequently, in step  48 , estimator  22  builds a first estimated channel impulse response using only the scatter pilots present in the symbols outputted by transformer  14 . The first estimated channel impulse response represents the channel power characteristic in the time domain in response to a predetermined impulse. Typically, the channel impulse response is computed using an IFFT (Inverse Fast Fourier Transform) within an IFFT-window. The IFFT-window is m· 
               T   u     n         
wide, where T u  is the duration of the modulation of an OFDM symbol that corresponds to the duration of an OFDM symbol minus the guard interval.  m  and  n  are the integer number previously defined.
 
       FIG. 3  shows an example of an estimated channel impulse response built by estimator  22  during step  48 . 
     The estimated channel impulse response has three power peaks P 1 , P 2  and P 3  within the IFFT-window. P 1  is the highest peak and corresponds to the main peak. 
     In the following part of the specification, we assume that P 2  is a real echo peak whereas P 3  is an aliased echo peak. In fact, peak P 3  corresponds to a real peak P 4  which is received just before the beginning of the IFFT-window. Thus, real echo peak P 4  is not visible in the estimated channel impulse response outputted by estimator  22 . 
     Next, we assume that the estimated channel impulse response built during step  48  is the one shown in  FIG. 3 . 
     The estimated channel impulse response is outputted to discriminator  30 . 
     In step  50 , discriminator  30  detects the position of the highest peak P 1 . Then, in step  52 , discriminator  30  detects the position of the other peaks. For example, discriminator  30  considers that there exists a peak if this peak is higher than a threshold S 1 . For example, threshold S 1  is set to be equal to A/q, where A is the amplitude of the main peak detected in step  50  and  q  is positive non zero integer. For example,  q  is greater than 4 and smaller than 128. In this embodiment,  q  is equal to 128. 
     Thereafter, in step  54 , discriminator  30  determines a time interval T i  between main peak P 1  and each of the other peaks within the IFFT-window. 
     Here, time interval T 1  refers to the time interval between peak P 1  and P 3  and time interval T 2  refers to the time interval between peak P 1  and peak P 2 . 
     In step  56 , discriminator  30  compares time intervals T 1  and T 2  with a predetermined threshold S 2 . If a time interval is smaller than or equal to threshold S 2 , then, in step  58 , discriminator  30  decides that the corresponding echo peak is not an aliased echo peak without any further computation. This results from the fact that it is highly probable that an echo peak, which is very close from the main peak, is a real echo peak. 
     For example, threshold S 2  is smaller than or equal to T u /2n. 
     In contrast, for each echo peak which is spaced apart from the main peak by more than S 2 , then: 
     in step  60 , discriminator  30  computes the correlation of N successive samples of the received digital data with a copy of the same N samples delayed by a time delay Δ 1,0  equal to time interval T 1 , and 
     in step  62 , discriminator  30  computes the correlation of the N successive samples of the received digital data with a copy of the same N samples delayed by a time delay Δ 2,0  equal to T IFFT -Δ 1,0 , where T IFFT  is the IFFT-window size. 
     For instance, N is chosen so that N·T e  is at least greater than T u  and preferably greater than 4·T u  or even greater than 16·T u . 
     More precisely, in step  60 , discriminator  30  computes other auto-correlation with a time delay Δ 1,i . Time delays Δ 1,i  is computed according to the following relation: Δ 1,i =T 1 +i·T e , where “i” is an integer that can be positive or negative. 
     Typically, the absolute value of “i” is never greater than N/100. 
     For instance, in this embodiment, only two other time delays Δ 1,−1  and Δ 1,1  are used. 
     Accordingly, step  60  includes three autocorrelation operations  65  to  67  using time delays Δ 1,−1 , Δ 1,0  and Δ 1,1 , respectively. Each one of these operations  65  to  67  is carried out in parallel by a corresponding correlator  32   a  to  32   c . 
     Similarly, step  62  includes three autocorrelation operations  70  to  72  using time delay T IFFT -Δ 1,−1 ; T IFFT -Δ 1,0  and T IFFT -Δ 1,1 . Operations  70  to  72  are carried out in parallel by corresponding auto-correlators  34   a  to  34   c . 
     Once the autocorrelation operations  65  to  67  have been carried out, in step  74 , discriminator averages or accumulates the autocorrelation results to obtain a global autocorrelation result C 1  based on each one of the autocorrelation results. 
     Similarly, once the autocorrelation operations  70  to  72  have been carried out, in step  76 , a global autocorrelation result C 2  is obtained by averaging or accumulating each one of the autocorrelation results from operation  70  to  72 . 
     Subsequently, in step  80 , discriminator  30  compares result C 1  with result C 2 . 
     If result C 1  is greater than result C 2 , then in step  82 , discriminator  30  decides that peak P 3  is a real echo peak so that there is no need to cancel this real echo peak from the estimated channel impulse response. 
     In contrast, if result C 1  is smaller than result C 2 , in step  84 , discriminator  30  decides that peak P 3  is an aliased echo peak. 
     Subsequently, in step  86 , discriminator  30  controls estimator  22  to cancel peak P 3  from the estimated channel impulse response. Preferably, estimator  22  replaces peak P 3  by an identical peak P 4  which is offset from peak P 3  by a time interval equal to T IFFT  either in the left direction or in the right direction. In fact, it is assumed that offsetting peaks P 3  in the right direction results in an echo peak at position e 1  and that offsetting peak P 3  in the left direction results in an echo peak at position e 2 . To decide in which direction peak P 3  should be offset, in this embodiment, corrector  24  always chooses the position e 1  or e 2  which is the closest from the main peak position. 
     Accordingly, in  FIG. 3 , peak P 3  is offset in the left direction and replaced by peak P 4 . 
     Steps  56  to  86  are iterated for each echo peak within the IFFT-window of the estimated channel impulse response to obtain a corrected channel impulse response. Once this has been done for every one of the echo peaks, the corrected estimated channel impulse is outputted to module  18  and equalizer  20 . Then, in step  90 , module  18  tunes the position of the FFT-window according to the corrected channel impulse response. For example, module  18  uses the position of peak P 1 . 
     In parallel, in step  92 , equalizer  20  uses the corrected estimated channel impulse response to tune its own coefficients. Subsequently, equalizer  20  equalizes the output of transformer  14  to correct errors due to disruption of the channel through which are received these data. 
       FIG. 4A  shows the analogic signal received through a main path.  FIG. 4B  shows an echo of this same signal received through a secondary path, which is longer than the main path. As a result, the echo is delayed by a time interval T in comparison to the signal received through the main path. In such a situation, the real channel impulse response has a real echo peak which is delayed by T from the main peak. We also assume that there is no other echo. 
     Thus, the correlation result of the received signal by a copy of the received signal delayed by a time delay different from T will result in a value close to zero because the transmitted data in the signal are supposed not to be correlated in the time domain. On the other hand, the correlation result of the received signal by a copy of the received signal delayed by time delay T will result in a value greatly different from zero. For instance, in this situation, the result of the correlation of the received signal by a copy of the received signal delayed by T IFFT -T will result in a value nearly null. As a consequence, an aliased echo peak can be discriminated from a real echo peak. 
     Many other embodiments are possible. For example, only one correlation is carried out in step  60  and  62 . Preferably, the only one correlation will be the correlation using time delay Δ 1,0 . This simplifies the method of  FIG. 2  while still obtaining good results for receiver implemented in a non-mobile terminal. A terminal is considered as non-mobile if it does not move at a speed greater than 10 Km/h. 
     In a very simple embodiment, either step  60  or step  62  is cancelled. Then, in step  80 , the global result C 1  or C 2  is compared to a predetermined threshold S 3  to discriminate between a real echo peak and an aliased echo peak. 
     In another embodiment, the N samples used to compute the correlation are not necessarily successive. 
     Receiver  6  and the corresponding method have been described in the particular case of an OFDM receiver. However, the above teaching can be applied in any telecommunication system in which the estimation of the channel impulse response may include aliased echo peaks due to sub-sampling. For example, the above method can be used in WIFI telecommunication systems.