Abstract:
A heat-sensitive type flow sensor which can be manufactured by using discrete parts at low cost in a miniature size with high accuracy and reliability includes a correcting circuitry capable of compensating for a detected current indicating a flow rate of a fluid of concern and outputted from a flow rate detecting circuit. In the heat-sensitive type flow sensor, an arithmetic processing circuit ( 200 ) for processing a detection voltage (V M ) supplied from a flow rate detecting circuit ( 100 ) and indicating the flow rate includes a voltage-to-current conversion circuitry ( 504 ) for converting the detection voltage (V M ) into a current (I 1 ) and capable of performing gain adjustment, a correcting circuitry ( 300 ) for correcting a level of a current signal (I 2 ) outputted from the voltage-to-current conversion circuit ( 500 ), and a detection current signal output circuit ( 600 ) for outputting the current signal (I 01 ) corrected by the correcting circuitry ( 300 ) as a detection output current signal of the heat-sensitive type flow sensor. The correcting circuitry ( 300 ) includes diodes implemented as discrete parts which need not exhibit identical characteristic.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a heat-sensitive type flow sensor for detecting a flow rate of a fluid by using heat-sensitive resistors. 
     2. Description of Related Art 
     In a conventional heat-sensitive type flow sensor known heretofore, a bridge circuit is employed which is constituted by a plurality of resistance elements inclusive of a first heat-sensitive resistor for detecting an atmospheric or ambient temperature and a second heat-sensitive resistor which is disposed within a passage through which a fluid flows such as, for example, an intake pipe of an internal combustion engine and heated electrically. To this end, a heating current supplied to the second heat-sensitive resistor for electrically heating it is so controlled that the temperature thereof is held higher than the ambient temperature by a predetermined value, wherein the quantity of heat which is deprived of the heat-sensitive resistor by the fluid whose flow rate is to be measured is detected in terms of change of the heating current flowing through the second heat-sensitive resistor. Thus, the flow rate of the fluid such as the intake air can be detected on the basis of the change of the heating current as detected. 
     For having better understanding of the present invention, description will first be made in some detail of a conventional heat-sensitive type flow sensor. FIG. 5 is a circuit diagram showing a circuit configuration of a hitherto known heat-sensitive type flow sensor such as disclosed, for example, in Japanese Unexamined Patent Application Publication No. 117436/1995 (JP-A-7-117436). Referring to the figure, the conventional heat-sensitive type flow sensor is comprised of a temperature control circuit  10 , an operational amplifier circuit  15 , a first constant current circuit  16 , a second constant current circuit  17 , a first constant current control circuit  18 , a third constant current circuit  19 , a fourth constant current circuit  20  and a second constant current control circuit  37 A, details of which will be described below. 
     At first, description will be directed to the temperature control circuit  10 . As can be seen in FIG. 5, the temperature control circuit  10  includes a bridge circuit constituted by resistors R 1  and R 2 , a flow rate detecting heat-sensitive resistor R h  and an ambient temperature detecting heat-sensitive resistor R c , wherein a junction f between the resistor R 1  and the ambient temperature detecting heat-sensitive resistor R c  and a junction b between the resistor R 2  and the flow rate detecting heat-sensitive resistor R h  of the bridge circuit are connected, respectively, to input terminals of a differential amplifier  101  which has an output terminal connected to the base of a transistor  102 , the emitter of which is connected to a junction a between the flow rate detecting heat-sensitive resistor R h  and the ambient temperature detecting heat-sensitive resistor R c  while the collector of the transistor  102  is connected to a positive or plus electrode of a DC power source  103  having the other electrode connected to the ground potential. 
     Next, description will turn to operation of the temperature control circuit  10 . When the voltages at the junctions b and f become equal to each other, the bridge circuit assumes an equilibrium or balanced state. In this state, an electric current I h  corresponding to the flow rate of a fluid concerned flows through the flow rate detecting heat-sensitive resistor R h . The output voltage V H  at the junction b can be given by a product of the current I h  and the resistance value of the resistor R 2 . This voltage V H  is made use of as a flow rate signal. 
     With the view to compensating for dispersions in the flow-rate detection due to dispersions of resistance values of the heat-sensitive resistors R h  and R c  and the resistors R 1  and R 2  as well as temperature coefficients thereof, the detection output value at a predetermined flow rate (ordinarily a relatively low flow rate) is set as a target or desired value by adjusting the resistance value of the resistor R 1  for thereby changing or translating the detection characteristic of the heat-sensitive type flow sensor correspondingly. 
     Description will now turn to the operational amplifier circuit  15  which is designed to process the flow rate signal outputted from the temperature control circuit  10 . The operational amplifier circuit  15  includes an operational amplifier  106  having an inverting input terminal and an output terminal interconnected by way of a feedback resistor R 13  and an input resistor R 11  having one end connected to the junction b of the bridge circuit mentioned above. The other end of the input resistor R 11  is connected to the non-inverting input terminal of the operational amplifier  106 . 
     The first constant current circuit  16  includes a transistor  110  having an emitter coupled to a line of a reference source voltage V ref  by way of a resistor R 25  and a collector connected to the non-inverting input terminal of the operational amplifier  106 . On the other hand, the second constant current circuit  17  includes a transistor  111  having an emitter electrode connected to the line of the reference source voltage V ref  by way of a resistor R 26  and a collector connected to the inverting input terminal of the operational amplifier  106 . The base electrodes of both the transistors  110  and  111  are connected in cascade and connected in common to an output terminal of an operational amplifier  108  which constitutes a part of the first constant current control circuit  18  which will be described below. 
     The first constant current control circuit  18  mentioned above is so designed as to control the output current values I 16  and I 17  of the first and second constant current circuits  16  and  17 , respectively, on the basis of the preset reference source voltage V ref . To this end, the first constant current control circuit  18  is constituted by resistors R 20 , R 21 , R 22 , R 23  and R 24  and an operational amplifier  108  connected in such a manner as can be seen in FIG.  5 . 
     Further, the heat-sensitive type flow sensor includes the third constant current circuit  19  of a structure similar to that of the first constant current circuit  16 , the fourth constant current circuit  20  implemented in an essentially same structure as that of the second constant current circuit  17  and the second constant current control circuit  37 A implemented similarly to the first constant current control circuit  18 . 
     Next, description will turn to operation of the operational amplifier circuit  15 . The value or voltage level of the input voltage V p  applied to the non-inverting input terminal of the operational amplifier  106  can be determined by subtracting from the output voltage V H  of the temperature control circuit  10  a voltage drop making appearance across the resistor R 11  due to the current I 16  flowing through the resistor R 11  by way of the first constant current circuit  16 . Namely, the input voltage V p  mentioned above can be given by the following expression (1): 
     
       
           V   p   =V   H −( R   11   ×I   16 )  (1)  
       
     
     On the other hand, the value or voltage level of the input voltage V n  applied to the inverting input terminal of the operational amplifier  106  can be determined by subtracting from the output voltage V o  of the operational amplifier circuit  15  a voltage drop making appearance across the resistor R 13  due to the current I 17  flowing through the resistor R 13  by way of the second constant current circuit  17 . Namely, the input voltage V p  mentioned above can be given by the following expression (2): 
     
       
           V   n   =V   o −( R   13   ×I   17 )  (2)  
       
     
     The operational amplifier  106  controls the output voltage V o  of the operational amplifier circuit  15  such that the condition given by V p =V n  is satisfied. Thus, the output voltage V o  of the heat-sensitive type flow sensor can be given by the following expression (3): 
     
       
           V   o   =V   H −( R   11   ×I   16 )+( R   13   ×I   17 )  (3)  
       
     
     In conjunction with the above expression (3), it is noted that when the resistance values of the resistors R 11  and R 13  are selected appropriately, e.g. R 11 =R 13 , the output voltage V o  of the operational amplifier circuit  15  can be expressed as follows: 
     
       
           V   o   =V   H −( I   17   +I   16 )×R 11   (4)  
       
     
     Next, operations of the first constant current circuit  16 , the second constant current circuit  17  and the first constant current control circuit  18  will be described. Since the input voltage V 20  applied to the non-inverting input terminal of the operational amplifier  108  is derived from the voltage division of the output voltage V H  of the temperature control circuit  10  by the resistors R 20  and R 21 , the input voltage V 20  is given by the following expression (5): 
     
       
           V   20   ={R   21 /( R   20   +R   21 )}× V   H   (5)  
       
     
     On the other hand, the input voltage V 22  applied to the inverting input terminal of the operational amplifier  108  is derived through voltage division of a potential difference between an inter-terminal voltage V 24  appearing across the resistor R 26  and the preset reference source voltage V ref  by the resistors R 22  and R 23 . Accordingly, the input voltage V 22  can be determined in accordance with the following expression (6): 
     
       
           V   22   =R   22 /( R   22   +R   23 )× V   24   +R   23 /( R   22   +R   23 )× V   ref   (6)  
       
     
     The operational amplifier  108  controls the first and second constant currents I 16  and I 17  flowing through the resistor R 24  by controlling the base currents of the transistors  110  and  111 , respectively, and further controls the inter-terminal voltage V 24  of the resistor  24  so that the condition V 20 =V 22  can be satisfied. Accordingly, the following expression (7) holds true. 
     
       
           R   21 /( R   20   +R   21 )× V   H   =R   22 /( R   22   +R   23 )× V   24   +R   23 /( R   22   +R   23 )× V   ref   (7)  
       
     
     From the expression (7), the inter-terminal voltage V 24  of the resistor R 24 , i.e., voltage appearing across the resistor R 24 , can be determined in accordance with the following expression (8): 
     
       
           V   24   =R   21 /( R   20   +R   21 )×( R   22   +R   23 )/ R   22   ×V   H   −R   22   /R   23   ×V   ref   (8)  
       
     
     In conjunction with the above expression (8), it is noted that when the resistances of the resistors R 20 , R 21 , R 22  and R 23  are dimensioned appropriately, e.g. R 20 =R 21  and R 22 =R 23 , then the expression (8) can be simplified as follows: 
     
       
           V   24   =V   H   −V   ref   (9)  
       
     
     Since the sum of the first and second constant currents, i.e., I 16 +I 17 , is equal to a sum of the current I 24  flowing through the resistor R 24  and the current I 23  flowing through a series circuit of the resistors R 23  and R 22 , i.e., I 24 +I 23 , the sum of the first and second constant currents, i.e., I 16 +I 17 , is given by the following expression (10): 
     
       
           I   16   +I   17   =V   24   /R   24 +( V   24   −V   ref )/( R   22   +R   23 )  (10)  
       
     
     In this conjunction, the following relations apply valid. 
     
       
           V   24   =V   H   −V   ref  and  R   22   =R   23   (11)  
       
     
     Accordingly, the sum of the first and second constant currents, i.e., I 16 +I 17 , is given by the following expression (12): 
     
       
           I   16   +I   17 =(1/ R   24 +1/2 R   22 )× V   H −(1/ R   24 +1/ R   22 )× V   ref   (12)  
       
     
     With the circuit configuration shown in FIG. 5, only the source current originating in the base currents of the transistors  110  and  111  is of significance to the output of the operational amplifier  108  without being accompanied by any sink current. Further, since the source current mentioned above becomes zero so long as the input voltage V 20  applied to the non-inverting input terminal of the operational amplifier  108  and the input voltage V 22  applied to the inverting input terminal thereof satisfy the condition that V 22 ≧V 20 , the first constant current I 16  as well as the second constant current I 17  is zero so long as the condition that V 22 ≧V 20  is satisfied. 
     Accordingly, the sum of the first and second constant currents, i.e., I 16 +I 17 , becomes zero on the conditions that R 20 =R 21  and R 22 =R 23 . Thus, the following expression (13) holds true: 
     
       
           V   24   +V   ref   ≧V   H   (13)  
       
     
     In that case, from the expression (10) and in view of the condition that R 22 =R 23 , the voltage V 24  can be given by the following expression (14): 
     
       
           V   24   ={R   24 /(2 R   22   +R   24 )}× V   ref   (14)  
       
     
     The inter-terminal voltage V 24  can not assume voltage level which is not higher than the value given by the above expression. Accordingly, on the condition given by the expression (15), the following expression (16) holds true. 
     
       
         {1 +R   24 /(2 R   22   +R   24 )}× V   ref   ≧V   H   (15),  
       
     
     
       
         and  
       
     
     
       
           I   16   +I   17 =0  (16)  
       
     
     Next, ratios or relation between the first constant current I 16  and the second constant current I 17  will be considered. Both the bases of the transistors  110  and  111  are at a same voltage level because they are connected in common to the output terminal of the operational amplifier  108 , as mentioned previously. Accordingly, assuming that the base-emitter voltage V be  of the transistor  110  and that of the transistor  111  are equal to each other, the inter-terminal voltage V 25  appearing across the resistor R 25  is equal to the inter-terminal voltage V 26  which makes appearance across the resistor R 26 . Thus, the relations given by the following expressions (17) applies valid. 
     
       
         
           V 
           25 
           =I 
           16 
           ×R 
           25  
         
       
     
     
       
         
           V 
           26 
           =I 
           17 
           ×R 
           26  
         
       
     
     
       
         V 25 =V 26   (17)  
       
     
     From the expression (17), the relation given by the following expression (18) applies valid. 
     
       
           I   16 =( R   26   /R   25 )× I   17   (18)  
       
     
     From the expressions (12) and (18), difference between the first constant current and the second constant current can be determined in accordance with the following expression (20):                        I   17     -     I   16       =                    (     1   -     G   1       )     /     (     1   +     G   1       )       ×     (       I   16     +     I   17       )                   =                    (     1   -     G   1       )     /     (     1   +     G   1       )       ×     {         (       1   /     R   24       +       1   /   2          R   22         )     ×     V   H       -                                    (       1   /     R   24       +     1   /     R   22         )     ×     V   ref       }                 (   20   )                                
     where G 1  represents R 26 /R 25 . 
     From the above expression (20) and the expression (4) concerning the output voltage V o  of the operational amplifier circuit  15 , relation given by the following expression (21) can apply valid.                      V   o     =                  V   H     +       R   11     ×       (     1   -     G   1       )     /     (     1   +     G   1       )       ×     (       I   16     +     I   17       )                     =                  V   H     +       R   11     ×       (     1   -     G   1       )     /     (     1   +     G   1       )       ×                                {         (       1   /     R   24       +       1   /   2          R   22         )     ×     V   H       -       (       1   /     R   24       +     1   /     R   22         )     ×     V   ref         }                   (   21   )                                
     As is apparent from the above expression (21), the output voltage V o  of the operational amplifier circuit  15  depends on the output voltage V H  of the temperature control circuit  10  and becomes equal to the output voltage V H  when the condition given by the following expression (22) is satisfied. 
     
       
           V   H ≦{1 +R   24 /(2 R   22   +R   24 )}× V   ref   (22)  
       
     
     Further, when the condition given by the following expression (23): 
     
       
           V   H &gt;{1 +R   24 /(2 R   22   +R   24 )}× V   ref   (23)  
       
     
     is satisfied, a value which depends on the difference between the output voltage V H  and the preset reference source voltage V ref , the resistance values of the resistors R 11 , R 22  and R 24  and the values of the ratio G 1  between the resistors R 25  and R 26  are added to or subtracted from the output voltage V H  in dependence on the magnitude or value of the above-mentioned ratio G 1  with reference to zero. 
     As will be appreciated from a foregoing description, in the conventional heat-sensitive type flow sensor, the circuit constants for the constant current control circuits are determined on the presumption that the base-emitter voltage V be  of the transistors  110  and  111  constituting the constant current circuits  16  and  17 , respectively, are equal to each other and that the terminal voltages appearing across the resistors connected to the emitters of the transistors, respectively, are equal to each other. 
     However, in many of the heat-sensitive type flow sensors known heretofore, the transistors  110  and  111  constituting parts of the constant current circuits  16  and  17 , respectively, are mounted as discrete components. Consequently, it is practically very difficult or impossible to ensure same operation characteristics for these transistors. In other words, the characteristics of these transistors will unavoidably differ from one to another. As a consequence, the output voltage V o  of the operational amplifier circuit can not necessarily bear correspondence relation to the output voltage V H  derived from the temperature control circuit with a reasonably sufficient fidelity. In order to eliminate these inconveniences, it is required to implement the first and second constant current circuits as well as the third and fourth constant current circuits as the integrated circuits to thereby uniformize the characteristics of these transistors. However, attempt for implementing the constant current circuits in the integrated circuit will encounter another problem that the cost involved in manufacturing the heat-sensitive flowmeter increases unprofitably. 
     SUMMARY OF THE INVENTION 
     In the light of the state of the art described above, it is an object of the present invention to provide a heat-sensitive type flow sensor which can be manufactured with high fidelity or accuracy and reliability even when the circuit elements mentioned above are realized by employing discrete parts. 
     Another object of the present invention is to provide a heat-sensitive type flow sensor which can be manufactured in a small size at low cost. 
     In general, the present invention teaches to provide a heat-sensitive type flow sensor with a correcting or compensating circuit which is capable of correcting or compensating for a detection current indicating a flow rate of a fluid of concern and outputted from a flow detecting means with high accuracy. 
     Thus, in view of the above and other objects which will become apparent as the description proceeds, there is provided according to a general aspect of the present invention a heat-sensitive type flow sensor which includes a flow rate detecting means disposed in association with a passage through which a fluid flows for detecting a flow rate of the fluid to thereby output a voltage of a level indicative of the flow rate, a voltage-to-current conversion means for converting the output voltage of the flow rate detecting means into a current, a correcting means for correcting a level of a current signal outputted from the voltage-to-current conversion means as the level of the output voltage indicative of the flow rate changes or drifts, and a current outputting means for outputting a current signal corrected by the correcting means as a detection output current signal indicative of the flow rate of the fluid. 
     By virtue of the provision of the current correcting or compensating means mentioned above, there can be realized a heat-sensitive type flow sensor whose detection output current signal is substantially evaded from the influences of voltage changes or drifts taking place in the heat-sensitive type flow sensor due to dispersion or variance of the characteristics of the circuit elements such as mentioned previously. Besides, the heat-sensitive type flow sensor can be manufactured inexpensively with uniformized characteristic by using discrete circuit elements. 
     In a mode for carrying out the invention, the correcting means should preferably include a means for making variable a correction starting point in dependence on change of the output voltage indicative of the flow rate. 
     With the arrangement mentioned above, the current correction or compensation starting point can be easily changed, to an advantage. 
     In another mode for carrying out the invention, the correcting means should preferably be so designed as to include a means for changing over increasing rate of the level of the current signal outputted from the current outputting means starting from the correction starting point. 
     Owing to the arrangement mentioned above, the current characteristic as demanded can be realized with ease, to another advantage. 
     In yet another mode for carrying out the invention, the correcting means should be so designed that the correction starting point is set at least at one of a time point at which the level of the output voltage indicative of the flow rate reaches or exceeds a preset reference voltage value and a time point at which the level of the output voltage indicative of the flow rate becomes lower than a preset reference voltage value. 
     Owing to the arrangement mentioned above, the current correction or compensation starting point can be easily changed, to an advantage. 
     In still another mode for carrying out the invention, a plurality of the correction starting points may be set in conjunction with the correcting means. 
     With the arrangement mentioned above, the correction or compensation starting point as well as the correcting or compensating current as demanded can be set without difficulty, to an advantage. 
     In a further mode for carrying out the invention, an output circuit of each of the voltage-to-current conversion means and the current outputting means may be constituted by a transistor circuit of Darlington connection type. 
     Owing to the arrangement mentioned above, errors or deviations of the base current of the transistor due to dispersion of the values and the temperature characteristics among the component of parts of the heat-sensitive type flow sensor can be reduced, which in turn means that the detection output current signal can be obtained with high fidelity and accuracy. 
     In a yet further mode for carrying out the invention, the voltage-to-current conversion means should preferably include a gain adjustment means for adjusting the level of the detection output current signal. 
     With the arrangement mentioned above, the number of the component parts can be reduced. Further, the heat-sensitive type flow sensor can be implemented on a substrate of a much reduced size, which will ultimately lead to implementing of the heat-sensitive type flow sensor in a miniaturized structure at low cost, to another advantage. 
     The above and other objects, features and attendant advantages of the present invention will more easily be understood by reading the following description of the preferred embodiments thereof taken, only by way of example, in conjunction with the accompanying drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In the course of the description which follows, reference is made to the drawings, in which: 
     FIG. 1 is a circuit diagram showing a circuit configuration of a heat-sensitive type flow sensor according to a first embodiment of the present invention; 
     FIGS. 2A to  2 C are characteristic diagrams for illustrating operations of a correcting or compensating circuit incorporated in the heat-sensitive type flow sensor according to the first embodiment of the invention, wherein 
     FIG. 2A is a view for illustrating operation of the correcting circuit in one state thereof; 
     FIG. 2B is a view for illustrating operation of the correcting circuit in another state thereof; and 
     FIG. 2C is a view illustrating operation of the correcting circuit in still another state thereof; 
     FIG. 3 is a circuit diagram showing a circuit configuration of a heat-sensitive type flow sensor according to a second embodiment of the present invention; 
     FIG. 4 is a circuit diagram showing a circuit configuration of a heat-sensitive type flow sensor according to a third embodiment of the present invention; and 
     FIG. 5 is a circuit diagram showing a conventional heat-sensitive type flow sensor known heretofore. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     The present invention will be described in detail in conjunction with what is presently considered as preferred or typical embodiments thereof by reference to the drawings. 
     Embodiment 1 
     Now, description will be made of a first embodiment of the present invention by reference to FIG. 1 which is a circuit diagram showing a circuit configuration of the heat-sensitive type flow sensor according to the first embodiment of the invention. Referring to the figure, the heat-sensitive type flow sensor now under consideration is comprised of a temperature control circuit  100  and an arithmetic processing circuit  200  which in turn includes a compensating or correcting circuitry  300  and an output circuitry  400 , wherein the output circuitry  400  is comprised of a voltage-to-current converting circuit  500  and a detection current signal output circuit  600 . 
     In general, the heat-sensitive type flow sensor according to the instant embodiment of the invention is implemented such that a detection voltage V M  outputted from the temperature control circuit  100  and indicating the flow rate of a flowing fluid of concern undergoes voltage-to-current conversion by the voltage-to-current converting circuit  500 , wherein a current I 2  resulting from the aforementioned voltage-current conversion is corrected or compensated for by the compensating or correcting circuitry  300  for outputting an output current signal I 01  indicating or reflecting the flow rate with high fidelity through the detection current signal output circuit  600 . 
     At first, description will be made of the temperature control circuit  100 . As can be seen in FIG. 1, the temperature control circuit  100  is comprised of a bridge circuit including a flow rate detecting heat-sensitive resistor R H , an atmospheric or ambient temperature detecting heat-sensitive resistor R K , a flow rate detecting fixed resistor R M  and a temperature compensating fixed resistor R 1 , a differential amplifier  101  having input terminals connected to junctions b and f, respectively, of the bridge circuit, a transistor  102  and a DC power source  103 , as is in the case of the conventional heat-sensitive type flow sensor. 
     More specifically, the output terminal of the differential amplifier  101  is connected to the base electrode of the transistor  102 , the emitter of which is connected to a circuit point a of the bridge circuit, while the collector of the transistor  102  is connected to the positive or plus electrode of the DC power source  103 , the other electrode of which is connected to the ground potential. Incidentally, the DC power source  103  may be constituted by a battery such as mounted on a motor vehicle. Parenthetically, operation of the temperature control circuit  100  is known and has been described hereinbefore in conjunction with the conventional heat-sensitive type flow sensor. Accordingly, repeated description thereof will be unnecessary. 
     Next, description will be directed to the arithmetic processing circuit  200 . The arithmetic processing circuit  200  is comprised of the output circuitry  400  designed to output the current I 2  through voltage-to-current conversion of the detection voltage V M  indicating the flow rate as supplied from the temperature control circuit  100  and the correcting circuitry  300  for correcting or compensating for the current I 2  resulting from the voltage-to-current conversion. 
     More specifically, the output circuitry  400  includes a voltage-to-current converting circuit  500  for converting the detection voltage V M  supplied from the temperature control circuit  100  and indicating the flow rate into a current I 2 , and a detection current signal output circuit  600  for outputting as a detection output current signal I 01  of the heat-sensitive type flow sensor the current I 2  resulting from the above-mentioned voltage-to-current conversion and indicating the flow rate after the correction or compensation of the current I 2  by the correcting circuitry  300 . 
     The voltage-to-current converting circuit  500  includes an operational amplifier  501  having an non-inverting input terminal to which the detection voltage V M  is applied by way of a fixed resistor  503 . A voltage-to-current converting resistor  504  is inserted between the inverting input terminal of the operational amplifier  501  and the ground potential. Further, the output terminal of the operational amplifier  501  is connected to the base of a transistor  505  by way of a fixed resistor  502 . On the other hand, the emitter of the transistor  505  is connected to the inverting input terminal of the operational amplifier  501 , while fixed resistors  605  and  606  constituting parts of the detection current signal output circuit  600  are connected in series between the collector of the transistor  505  and a terminal of the reference source voltage V cc . 
     Next, description will turn to operation of the voltage-to-current converting circuit  500 . The operational amplifier  501 , the fixed resistor  502  and the transistor  505  cooperate to constitute a negative feedback circuit. Thus, representing by V M ′ the potential at the inverting input terminal of the operational amplifier  501 , relation given by the following expression (24) applies valid. 
     
       
         V M ′=V M   (24)  
       
     
     The potential or voltage V M ′ is converted into a current I 1  by means of the voltage-to-current converting resistor  504  in accordance with the following expression (25): 
     
       
         I 1 =V M ′/R 2 =V M /R 2   (25,  
       
     
     where R 2  represents the resistance value of the voltage-to-current converting resistor  504 . 
     In this manner, the detection voltage V M  indicative of the flow rate of a fluid undergoes the voltage-to-current conversion to be thereby converted into the detection current I 1  indicative of the flow rate. In this conjunction, it is to be mentioned that by implementing the voltage-to-current converting resistor  504  having the resistance value R 2  as an adjustable variable resistor instead of the fixed resistor, it is possible to effectuate both the voltage-to-current conversion and the gain adjustment concurrently. Representing the collector current and the emitter current of the transistor  505  by I c1  and I e1 , respectively, the following expression (26) applies valid. 
     
       
         I c1 ≈I e1 =I 1   (26)  
       
     
     Turning to the detection current signal output circuit  600 , it includes an operational amplifier  601  having a non-inverting input terminal which is connected to a junction between the fixed resistors  605  and  606  with the inverting input terminal thereof being connected to the terminal of the reference source voltage V cc  by way of a fixed resistor  604 . Further, the output terminal of the operational amplifier  601  is connected to the base of a transistor  603  by way of a fixed resistor  602 . On the other hand, the emitter of the transistor  603  is connected to the inverting input terminal of the operational amplifier  601 , wherein the detection output current signal I 01  of the heat-sensitive type flow sensor is outputted from the collector of the transistor  603 . 
     Operation of the detection current signal output circuit  600  will now be described without taking into consideration the operation of the correcting circuitry  300  for simplification of the description. Owing to the voltage-to-current conversion performed by the voltage-to-current converting circuit  500 , the detection current I 1  indicative of the flow rate of a fluid flows through the transistor  505 , the fixed resistor  605  having a resistance value R 3  and the fixed resistor  606  having a resistance value R 4 . In that case, the inter-terminal voltages V 1  and V 2  making appearance across the fixed resistors  605  and  606 , respectively, can be represented by the following expressions (27) and (28) on the basis of the expression (25). 
     
       
           V   1   =V   cc −{( R   3   +R   4 )/ R   2   }×V   M   (27)  
       
     
     
       
           V   2   =V   cc −( R   3   /R   2 ) ×V   M   (28)  
       
     
     Since the operational amplifier  601 , the fixed resistor  602  and the transistor  603  cooperate to constitute the negative feedback circuit, as mentioned preciously, a so-called imaginary short-circuit state prevails. Consequently, when the potential at the inverting input terminal of the operational amplifier  601  is represented by V 2 ′, the following expression (29) applies valid. 
     
       
         V 2 ′=V 2   (29)  
       
     
     Thus, the current I o  given by the undermentioned expression (30) flows through a fixed resistor  604  having a resistance value R 5 . 
     
       
           I   o =( V   cc   −V   2 )/ R   5   ={R   3 /( R   2   ×R   5 )}×V M   (30)  
       
     
     Representing the collector current and the emitter current of the transistor  603  by I c2  and I e2 , respectively, the following expression (31) applies valid. 
     
       
         I c2 ≈I e2 =I o   (31)  
       
     
     In reality, however, the current flowing through the fixed resistors  605  (R 3 ) and  606  (R 4 ) is given as a sum of the detection current I 1  indicating the flow rate and the current I 2  because the correcting circuitry  300  is provided. 
     The compensating or correcting circuitry  300  is constituted by operational amplifiers  301  and  306 , diodes  302  and  307 , fixed resistors  303 ,  304 ,  308  and  309 , and variable resistors  305  and  310  which are interconnected in such a manner as can be seen in the figure. 
     In operation, it is assumed that the correcting current I 2  flows toward the correcting circuitry  300 , as illustrated in FIG.  1 . Then, the voltages V 1  and V 2  making appearance across the fixed resistors  605  and  606 , respectively, are given by the following expressions (32) and (33), respectively. 
     
       
           V   1   =V   cc −( R   3   +R   4 )×( I   1   +I   2 )  (32)  
       
     
     
       
           V   2   =V   cc   −R   3 ×( I   1   +I   2 )  (33)  
       
     
     where I 1  represents the detection current depending on the flow rate, and I 2  represents the current component utilized for the correction and is referred to as the compensating or correcting current. 
     From the expressions (30) and (33), the current I o  flowing through the fixed resistor  604  is given by the following expression (34): 
     
       
           I   o =( R   3   /R   5 )×( I   1   +I   2 )  (34)  
       
     
     At this juncture, the potentials or voltages V r1  and V r2  at the non-inverting input terminals of the operational amplifiers  301  and  306 , respectively, can be given by the following expressions (35) and (36), respectively. 
     
       
           V   r1   ={R   7 /( R   6   +R   7 )}× V   cc   (35)  
       
     
     
       
           V   r2   ={R   9 /( R   8   +R   9 )}× V   cc   (36)  
       
     
     Because of the ideal diode connections, respectively, between the operational amplifier  301  and the diode  302  and between the operational amplifier  306  and the diode  307 , as shown in FIG. 1, such correcting current I 2  will flow to the correcting circuitry  300  which satisfies the conditions given by the following expressions (37) and (38): 
     
       
         V r1 =V r1 ′  (37)  
       
     
     
       
         V r2 =V r2 ′  (38)  
       
     
     where V r1 ′ and V r2 ′ represent potentials at the inverting input terminals of the operational amplifiers  301  and  306 , respectively. 
     Correction or compensation performed by the correcting circuitry  300  will be elucidated by referring to FIGS. 2A to  2 C. FIG. 2A is a view illustrating graphically V M -versus-V 1  characteristic and V M -versus-I o  characteristic on the assumption that a variable resistor  310  having a resistance value R 11  shown in FIG. 1 is opened. By taking as a parameter the detection voltage V M  outputted from the temperature control circuit  100 , the potential V 1  and the current I o  (see FIG. 1) can be expressed as follows: 
     
       
           V   1   =V   cc −{( R   3   +R   4 )/ R   2   }×V   M    
       
     
     
       
           I   o   ={R   3 /( R   2   ×R   5 )}×V M    
       
     
     The correction of the detection current I 1  is performed with the non-inverting input voltage V r1  set in accordance with the expression (35) in such a manner as described below. 
     When V 1 &gt;V r1 , the direction in which the correcting current I 2  flows is reverse to the forward direction of the diode  302 . Accordingly, no correcting current can flow. By contrast, when V 1 ≦V r1 , as shown in FIG. 2A, the direction of the correcting current I 2  coincides with the forward direction of the diode  302 . Accordingly, the correcting current I 2  flows. 
     Thus, the condition enabling the correction or compensation of the detection current is given by 
     
       
         V 1 ≦V r1   (39)  
       
     
     where V r1  is given by 
     
       
           V   r1   =V   cc −{( R   3   +R   4 )/ R   2   }×V   M   (40)  
       
     
     Representing by V M1  the detection voltage V M  which satisfies the condition given by the expression (40), the condition enabling the correction or compensation can be expressed as follows: 
     
       
         V M &gt;V M1   (41)  
       
     
     On the other hand, the correcting current I 2  can be determined as follows: 
     
       
           I   2 =( V   1   −V   r1 )/ R   10   (42)  
       
     
     From the expressions (25), (42) and (34), the current I o  can be given by the following expression (43): 
     
       
           I   o   ={R   3 /( R   2   ×R   5 )} V   M   +{R   3 /( R   5   ×R   10 )}×( V   1   −V   r1 )  (43)  
       
     
     On the basis of the expressions (35) and (43), the correcting point can easily be set by adjusting the ratio of voltage division effected by the fixed resistor  303  (R 6 ) and the fixed resistor  304  (R 7 ), while the correcting current I 2  can easily be set by means of the variable resistor  305  (R 10 ). 
     FIG. 2B is a view illustrating graphically V M -versus-V 1  characteristic and V M -versus-I o  characteristic on the assumption that the variable resistor  305  having a resistance value R 10  shown in FIG. 1 is opened. By taking the detection voltage V M  as a parameter, the potential V 1  and the current I o  can be expressed as follows: 
     
       
           V   1   =V   cc −{( R   3   +R   4 )/ R   2   }×V   M    
       
     
     
       
         I o   ={R   3 /( R   2   ×R   5 )}×V M    
       
     
     The correction of the detection current I 1  is performed with the non-inverting input voltage V r2  set in accordance with the expression (36) in the manner described below. 
     When V 1 &lt;V r2 , the direction in which the correcting current I 2  flows is reverse to the forward direction of the diode  307 . Accordingly, no correcting current can flow. By contrast, when V 1 ≧V r2 , as shown in FIG. 2B, the direction of the correcting current I 2  coincides with the forward direction of the diode  307 . Accordingly, the correcting current I 2  flows. 
     Thus, the condition enabling the correction of the detection current is given by 
     
       
         V 1 ≧V r2   (44)  
       
     
     where V r2  is given by 
     
       
           V   r2   =V   cc −{( R   3   +R   4 )/ R   2   }×V   M   (45)  
       
     
     Representing by V M2  the detection voltage V M  which satisfies the condition given by the expression (45), the condition enabling the correction can be expressed as follows: 
     
       
         V M ≦V M2   (46)  
       
     
     On the other hand, the correcting current I 2  can be determined as follows: 
     
       
           I   2 =( V   1   −V   r2 )/ R   11   (47)  
       
     
     From the expressions (25), (47) and (34), the current I o  can be given by the following expression (48): 
     
       
           I   o   ={R   3 /( R   2   ×R   5 )}×V M   +{R   3 /( R   5   ×R   11 )}×( V   1   −V   r2 )  (48)  
       
     
     From the expressions (36) and (48), the correcting point can easily be set by adjusting the ratio of voltage division effected by the fixed resistor  308  (R 8 ) and the fixed resistor  309  (R 9 ), while the correcting current I 2  can easily be set by means of the variable resistor  310  (R 11 ). 
     FIG. 2C is a diagram illustrating graphically V M -versus-V 1  characteristic and the V M -versus-I o  characteristic when both the corrections illustrated in FIGS. 2A and 2B are concurrently effectuated. In this case, the correcting point and the current for the correction or compensation of the detection signal can be easily set, as will be readily understood from the description made with reference to FIGS. 2A and 2B. 
     Embodiment 2 
     According to the teaching of the present invention incarnated in the second embodiment thereof, it is proposed to implement the transistor  505  and the transistor  603  which constitute parts of the voltage-to-current converting circuit  500  and the detection current signal output circuit  600 , respectively, in the form of Darlington transistor circuits  603 A and  505 A, respectively, as shown in FIG.  3 . With such circuit arrangement, there can be made available a large current amplification factor. This in turn means that errors due to dispersion or nonuniformity among the component parts as well as variations of the base currents due to temperature characteristics of the transistors can be suppressed, whereby the output signal of the heat-sensitive type flow sensor can be obtained with high accuracy and reliability. 
     Embodiment 3 
     In the heat-sensitive type flow sensor according to a third embodiment of the present invention, the electronic parts constituting the ideal diode circuits are implemented by using base-emitter junctions (PN-junctions) of transistors  302 A and  307 A. By virtue of this arrangement, it becomes unnecessary to use the diodes of identical characteristic. In other words, the ideal diode circuits can be implemented by using discrete parts, which in turn means that the cost for fabrication of the heat-sensitive type flow sensor can be reduced, to another advantage. 
     Many modifications and variations of the present invention are possible in the light of the above techniques. It is therefore to be understood that within the scope of the appended claims, the invention may be practiced otherwise than as specifically described.