Abstract:
A method includes energizing and de-energizing at least one inductor to furnish an output power. An indication of the output power is monitored to detect when a transient occurs in the output power. The energizing and de-energizing are performed at a predetermined frequency when the transient is not detected; and in response to the detection of the transient, the energizing and de-energizing are performed at a frequency that deviates from the predefined frequency to compensate for the transient.

Description:
BACKGROUND 
     The invention generally relates to a voltage regulator, such as a switching voltage regulator, that adjusts a timing in response to a load transient. 
     A DC-to-DC voltage regulator typically is used to convert a DC input voltage to either a higher or a lower DC output voltage. One type of voltage regulator is a switching regulator that is often chosen due to its small size and efficiency. The switching regulator typically includes one or more switches that are rapidly opened and closed to transfer energy between an inductor (a stand-alone inductor or a transformer, as examples) and an input voltage source in a manner that regulates the output voltage. 
     As an example, referring to FIG. 1, one type of switching regulator is a Buck switching regulator  10  that receives an input DC voltage (called V IN ) and converts the V IN  voltage to a lower regulated output voltage (called V OUT ) that appears at an output terminal  11 . To accomplish this, the regulator  10  may include a switch  20  (a metal-oxide-semiconductor field-effect-transistor (MOSFET), for example) that is operated (via a voltage called V SW ) in a manner to regulate the VOUT voltage, as described below. 
     Referring also FIGS. 2 and 3, in particular, the switch  20  opens and closes to control energization/de-energization cycles  19  (each having a constant duration called T S ) of an inductor  14 . In each cycle  19 , the regulator  10  asserts, (drives high, for example) the V SW  voltage during an on interval (called T ON ) to close the switch  20  and transfer energy from an input voltage source  9  to the inductor  14 . During the T ON  interval, a current (called I L ) of the inductor  14  has a positive slope. During an off interval (called T OFF ) of the cycle  19 , the regulator  10  deasserts (drives low, for example) the V SW  voltage to open the switch  20  and isolate the input voltage source  9  from the inductor  14 . At this point, the level of the I L  current is not abruptly halted, but rather, a diode  18  begins conducting to transfer energy from the inductor  14  to a bulk capacitor  16  and a load (not shown) that are coupled to the output terminal  11 . During the T OFF  interval, the I L  current has a negative slope, and the regulator  10  may close a switch  21  to shunt the diode  18  to reduce the amount of power that is otherwise dissipated by the diode  18 . The bulk capacitor  16  serves as a stored energy source that is depleted by the load, and additional energy is transferred from the inductor  14  to the bulk capacitor  16  during each T ON  interval. 
     For the Buck switching regulator, the ratio of the T ON  interval to the T OFF  interval, called a duty cycle, generally governs the ratio of the V OUT  to the V IN  voltages. Thus, to increase the V OUT  voltage, the duty cycle may be increased, and to decrease the V OUT  voltage, the duty cycle may be decreased. 
     As an example, the regulator  10  may include a controller  15  (see FIG. 1) that regulates the V OUT  voltage by using a fixed frequency, pulse width modulation (PWM) technique to control the duty cycle. In this manner, the controller  15  may include an error amplifier  23  that amplifies the difference between a reference voltage (called V REF ) and a voltage (called V P  (see FIG.  1 )) that is proportional to the V OUT  voltage. Referring also to FIG. 5, the controller  15  may include a comparator  26  that compares the resultant amplified voltage (called V C ) with a sawtooth voltage (called V SAW ) and provides the V SW  signal that indicates the result of the comparison. The V SAW  voltage is provided by a sawtooth oscillator  25  and may have a constant frequency (i.e., 1/T S ). 
     Due to the above-described arrangement, when the V OUT  voltage increases, the V C  voltage decreases and causes the duty cycle to decrease to counteract the increase in V OUT . Conversely, when the V OUT  voltage decreases, the V C  voltage increases and causes the duty cycle to increase to counteract the decrease in V OUT . The switching frequency (i.e., 1/T S ) typically controls the magnitude of an AC ripple component (called V RIPPLE  (see FIG.  4 )) of the V OUT  voltage, as a higher switching frequency typically reduces the magnitude of the V RIPPLE  voltage. 
     The regulator  10  may be part of a computer system and thus, may be used to provide power to components, such as a microprocessor, of the computer system. Because of the ever-increasing operating frequency and power requirements of the microprocessor, the microprocessor may consume a significant amount of power. When the power that is demanded by the microprocessor suddenly increases, giving rise to a transient condition, the voltage that is supplied by the regulator  10  may tend to decrease below an acceptable range of voltages. To prevent this from occurring, the computer system may include a significant amount of decoupling capacitors (not shown) to prevent the voltage that supplies the microprocessor from substantially decreasing when the output load of the regulator  10  suddenly changes. Without the decoupling capacitors, the voltage supplied to the microprocessor may drop below an acceptable level due to the above-described PWM control. In this manner, when a significant load transient occurs, the control scheme may be within a dead time interval, a time interval in which the switch  20  is open, thereby preventing energy from being transferred from the input source  9  to counteract the transient. In general, the response of the regulator  10  to a load transient is a function of the inductance of the inductor  14 . Although the current in the inductor  14  cannot change instantaneously when the switch  20  closes, in general, the smaller the inductance of the inductor  14 , the faster the regulator  10  may respond to counteract the transient. However, if the control scheme is within the dead interval, an additional time elapses in which the regulator  10  cannot respond to the transient. 
     For example, referring to FIGS. 6 and 7, the controller  15  (see FIG. 1) may generate pulse width modulated pulses  30  (via the V SW  signal) to regulate the V OUT  voltage for a given level (called I CC—MIN ) of output current (called I OUT ) of the regulator  10 . At time T 3 , the pulse  30   a  may end, thereby causing the switch  20  to open at time T 3 . However, also at time T 3 , the I OUT  current may transition from the I CC—MIN  level to a higher current level (called I CC—MAX ). The controller  15  may not close the switch  20  until another switching cycle begins (and until another pulse  30   b  is generated) at time T 4 . Therefore, a dead time interval  32  may occur in which the switch  20  is open, a state of the regulator  10  that prevents the regulator  10  from immediately responding to the increased load. 
     The duration of the dead interval  32  may be reduced by coupling two of the regulators  10  in parallel and operating their switches  20  in a complementary fashion. However, this arrangement may also not respond fast enough to prevent a significant drop in the regulator&#39;s output voltage. 
     Thus, there is a continuing need for a switching regulator having an improved response to load transients. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     FIG. 1 is a schematic diagram of a switching voltage regulator of the prior art. 
     FIGS. 2,  3 ,  4 ,  5 ,  6  and  7  are current and voltage waveforms that illustrate operation of the regulator of FIG.  1 . 
     FIG. 8 is a switching voltage regulator according to an embodiment of the invention. 
     FIGS. 9,  10 ,  11 ,  12 , and  13  depict waveforms illustrating operation of the regulator of FIG. 8 according to different embodiments of the invention. 
     FIG. 14 is a schematic diagram of a sawtooth oscillator of FIG. 8 according to an embodiment of the invention. 
     FIG. 15 depicts waveforms illustrating performance of the regulator according to different embodiments of the invention. 
     FIG. 16 is a schematic diagram of a computer system according to an embodiment of the invention. 
    
    
     DETAILED DESCRIPTION 
     Referring to FIG. 8, an embodiment  40  of a switching voltage regulator in accordance with the invention includes two Buck switching regulator stages  42   a  and  42   b  (each having a similar design denoted by the reference numeral “ 42 ”) that are coupled together in parallel to convert an input voltage (called “VIN”) into an output voltage (called V OUT ). In this manner, the stages  42   a  and  42   b  each receive the V IN  input voltage from a voltage source  52  and regulate the V OUT  voltage that appears at an output terminal  80  that is common to both of the stages  42   a  and  42   b . The stages  42   a  and  42   b  also share a controller  60  (as described below) and a bulk capacitor  53  that is coupled between the output terminal  80  and ground. 
     More particularly, in some embodiments of the invention, each stage  42  includes a switch  47  (a metal-oxide-semiconductor field-effect-transistor (MOSFET), for example) that is coupled between the positive terminal of the voltage source  52  and a terminal of an inductor  45 . The other terminal of the inductor  45  is coupled to the output terminal  80 . For the stage  42   a , the switch  47  is closed and opened by a signal called V SW1 , and for the stage  42   b , the switch  47  is closed and opened by a signal called V SW2 . For each stage  42 , the closing of the switch  47  causes energy to be transferred from the source  52  and stored in the inductor  45  to energize the inductor  45 , and the opening of the switch  47  causes the stored energy to be transferred from the inductor  45  to the output terminal  80 , a transfer that de-energizes the inductor  45 . 
     In some embodiments of the invention, the controller  60  generates the V SW1  and V SW2  signals in a manner that, in general, causes the on times (the times in which the switch  47  conducts, or is closed) of the two switches  47  to be shifted 180° apart. In other embodiments of the invention, the regulator may be a multiphase regulator other than a two phase regulator, and in these embodiments, the controller  60  may generate signals to control the operation of the stages so that the switch control signals have the proper phase relationship. As examples, for a three phase regulator, the switch control signals are 120° apart, for a four phase regulator, the switch control signals are 90° apart, etc. 
     For the two phase design (assumed in the description below unless otherwise noted), the V SW1  signal includes pulses  90  (see FIG.  9 ), each of which has a duration that sets the on time of a particular switching cycle of the stage  42   a  and appears at a predefined switching frequency. 
     Similarly, the V SW2  signal includes pulses  94  (see FIG.  10 ), each of which has a duration that sets the on time of a particular switching cycle of the stage  42   b  and appear at a predefined switching frequency. Furthermore, the pulses  94  are generally shifted 180° with respect to the pulses  90 . Assuming no transients occur, the controller  60  generates the pulses  90  at a predefined switching frequency; generates the pulses  94  at a predefined switching frequency; and regulates the V OUT  voltage by adjusting the duty cycles of the V SW1  and V SW2  signals. 
     Due to the phase shift between the pulses  90  and  94  that is introduced by the controller  60 , a ripple noise component of the V OUT  signal may be substantially minimized because the ripple currents that are contributed by each stage  42  are 180° out of phase. Furthermore, the frequency of the ripple noise component is twice the switching frequency of the stages  42 , thereby permitting the use of a smaller capacitor  53  than would otherwise be used without the frequency doubling. 
     Although the regulator  40  has an effective switching frequency that is approximately twice the switching frequency of either stage  42 , the regulator  40  may still not respond quickly enough to a sudden change (i.e., a transient) in the power that is demanded by a load of the regulator  40  if not for the features described below. For example, FIGS. 9,  10  and  13  depict a possible worst case scenario. In this scenario, at time T 0 , the output power of the regulator  40  may suddenly increase, as depicted by the increase in the I OUT  current level from an I CC—MIN  level to an I CC—MAX  level at time To in FIG.  13 . Time T 0  is also the time at which the on time of the pulse  90   a  (see FIG. 9) would normally end. Thus, a dead time interval  99  (see FIGS. 9 and 10) in which the switches  47  of both stages  42  are open may exist between the end of the pulse  90   a  and the beginning (time T 1 ) of the pulse  94   a , if not for the features described below. In a conventional regulator, the regulator may not close a switch to transfer energy from an input source to an inductor during the dead time interval, and thus, in the conventional regulator, additional energy may not be transferred from the input source to counteract the transient. 
     As described below, for purposes of preventing this from happening, for the scenario described above, the controller  60  begins another on time interval (to extend the on time interval  90   a  by the additional portion  93 ) at time T 0 . In this manner, the controller  60  monitors the output power (as described below) to detect a load transient. When the controller  60  detects a transient, the controller  60  deviates from the predetermined switching frequency to counteract the increase in load, a control technique that departs from the periodic schedule that the controller  60  otherwise follows to generate the V SW1  and V SW2  signals. Thus, due to this control technique, the switch  47  of the stage  42   a  has more on time to counteract the transient. As depicted by the example shown in FIG. 9, the on time of the pulse  90   a  is extended so that the pulse  90  includes a portion  92  due to the non-transient related control and a portion  93  that is attributable to the transient compensation. 
     Because the occurrence of a transient (such as the transient that is depicted in FIG. 13) may be viewed as a random event, the transient may occur after the occurrence of a pulse  90  and before the occurrence of a pulse  94  (as depicted in FIGS. 9 and 10) or after the occurrence of a pulse  94  and before the occurrence of the pulse  90 . However, regardless of the timing of the transient, the controller  60  alters the timing of the appropriate V SW1  or V SW2  signal to effectively add more on time to counteract the effect of the transient. Thus, if the controller  60  detects a transient after the expiration of one of the pulses  90  and before the next pulse  94 , the controller  60  may either (depending on the particular embodiment) create another pulse  90  or  94  ahead of schedule. If the transient occurs during one of the pulses  90  and  94 , in some embodiments, the controller  60  does not create additional on time for either switch  47 , as one of the switches  47  is closed when the transient occurs. However, in other embodiments, the controller  60  may extend the on time of one of the switches  47  if the transient occurs during one of the pulses  90  and  94 . 
     The control scheme that is described above assumes complementary switching of the stages  42   a  and  42   b , i.e., the switch  47  of one stage  42  is open while the switch  47  of the other stage  42  is closed. However, in some embodiments, the controller  60  temporarily switches the converters  42  in parallel to respond to the transient. For example, for the scenario that is depicted in FIGS. 9,  11  and  13 , the controller  60 , in some embodiments of the invention, may generate the pulse  94   a  in synchronization with the latter portion  93  of the pulse  90   a , beginning at time T 0 . Thus, due to this arrangement, the source  52  may communicate energy concurrently through both converters  42   a  and  42   b  to counteract the transient. As shown, after the pulses  90   a  and  94   a , the controller  60  returns to the above-described complementary switching constant frequency PWM control scheme. 
     Referring back to FIG. 8, among the other features of each Buck converter stage  42 , the stage  42  includes a diode  49  that has its cathode coupled to the inductor terminal that is closest to the switch  47 . The anode of the diode  49  is coupled to a current sensing resistor  52  that is coupled between the anode and ground. The resistor  52  of the stage  42   a  furnishes a voltage (called V 1 ) that indicates the inductor current of the stage  42   a , and the resistor  52  of the stage  42   b  furnishes a voltage (called V 2 ) that indicates the inductor current of the stage  42   b . In some embodiments of the invention, each stage  42  includes a switch  50  that is coupled in parallel with the diode  49  and may be used to reduce resistive power losses in the stages  42 . 
     In some embodiments of the invention, the controller  60  regulates the V OUT  voltage by using a constant frequency pulse width modulation (PWM) control technique to control the duty cycle of the two switches  47 , except when a transient occurs. The control described below is a voltage mode PWM control (except when a transient occurs). However, it is contemplated that in other embodiments of the invention, other control schemes (a current mode control scheme, for example) may be used to control operation of the switches  47  during times in which a transient does not occur. 
     For the voltage mode control (assumed in the description below unless otherwise noted), the controller  60  may include an error amplifier  68  that amplifies the difference between a reference voltage (called V REF ) and a voltage (called V P ) that is proportional to the V OUT  voltage. A comparator  64  of the controller  60  compares the resultant amplified voltage (called V C ) with a sawtooth voltage (called V SAW ) and provides a signal (to a control circuit  62 ) that indicates the result of the comparison. The V SAW  voltage is provided by a sawtooth oscillator  25  and has a constant switching frequency until reset by the controller  60  to cause a deviation from the constant frequency to counteract a transient, as described below. 
     More particularly, in some embodiments of the invention, the control circuit  62  may monitor the output power of the regulator  40  through the V 1  and V 2  voltages, each of which indicates the current in a different one of the inductors  45 . Of course, the control circuit  62  may use other techniques to monitor the output power. For example, the control circuit  62  may monitor currents through coils that are magnetically coupled to the inductors  45 , for example. Regardless of the technique used to monitor the power output of the regulator  40 , the control circuit  62  monitors the output power of the regulator  40  to detect a transient in the output power. As an example, the control circuit  62  may compare the measured output current to a predetermined threshold level to identify the occurrence of a transient. If a substantial change in the output current occurs during a predetermined time interval (a switching period, for example) then, in some embodiments, the control circuit  62  deems this to be a transient. A substantial change in the output current may be indicated by the output current surpassing, as examples, an absolute current level or a current level that is a predetermined amount above the output current level that existed before the beginning of the predetermined time interval. 
     When the regulator  40  does not encounter a transient, the control circuit  62  operates in the following manner. When the comparator  64  asserts (drives high, for example) its output signal, the control circuit  62  pulses either the V SW1  or the V SW2  signal high to maintain the interleaved switching of the stages  40   a  and  40   b . Thus, as depicted in FIG. 12, for each switching cycle, as long as the V C  voltage is greater than the V SAW  signal, the control circuit  62  asserts (drives high, for example) the V SW1  or V SW2  signal to generate the pulse  90 ,  94 . Otherwise, the control circuit  62  deasserts (drives low, for example) both the V SW1  and V SW2  signals. 
     When the control circuit  62  detects a transient, the control circuit  62  responds in the following manner. First, the control circuit  62  determines if one of the switches  47  is already closed. If so, then in some embodiments of the invention, the control circuit  62  takes no further action than the scheduled above-described PWM control, as energy is being communicated from the input source  52  to counteract the transient. Otherwise, if no switch  47  is closed when the transient occurs, the control circuit  62  resets the sawtooth oscillator  66 , as depicted in the V SAW  (see FIG. 12) signal at time T 0 . The control circuit&#39;s reset of the V SAW  signal creates another on time interval for the converter  42  to respond to the transient. In this manner, as depicted in FIG. 12, the V C  voltage may rise after time T 0  due to a decrease in the V OUT  voltage. However, by resetting the sawtooth oscillator  66 , another on time interval is available for one or both (depending on the particular embodiment) of the converters  42  to respond to the transient to keep the V OUT  voltage within regulation. 
     Referring to FIG. 14, in some embodiments of the invention, the sawtooth oscillator  66  includes a ramp generator, or integrator, that includes an integrating capacitor  108  that is coupled between ground and a constant current source  110 . The V SAW  signal is the voltage drop across the capacitor  108 . A switch  106  is coupled in parallel with the capacitor  108  and is operated by a pulse generator  102  and the control circuit  62  to create the V SAW  signal, as described below. 
     In this manner, the pulse generator  102  generates a periodic pulse train signal (called V PP ) that when de-asserted (driven low, for example) permits the V SAW  voltage to ramp upward. The V PP  signal sets the predefined switching and thus, has a frequency when not transient occurs near the frequency of the V SAW  frequency. When the V PP  signal is asserted (driven high, for example) due to one of the narrow pulses of the pulse train signal, the switch  106  closes to discharge the capacitor  108  and cause the V SAW  voltage to decrease to form one of the vertical edges of the V SAW  signal. 
     The switch  106  is controlled by the voltage that appears on the output terminal of an OR gate  103 . One input terminal of the OR gate  103  is connected to a reset line  109  that is coupled to from the control circuit  62 , and another input terminal of the OR gate  103  receives the V PP  signal. Thus, due to this arrangement, the V PP  signal controls the periodic timing of the V SAW  and thus, controls the periodic scheduling of the on times for the V SW1  and V SW2  signals when the regulator  40  is not responding to a transient condition. However, the control circuit  62  may reset the sawtooth generator  66  by asserting (driving high, for example) the reset line  109 , an action that closes the switch  106  and resets the ramp generator. 
     Referring to FIG. 15, the in phase and out of phase versions of the above-described circuit were simulated along with a conventional multi-phase parallel regulator that was formed from buck converter stages. The simulation included simulating a transient in the power demand. As shown by a waveform  122 , the constant timing approach of the conventional regulator permits a significant voltage drop to occur when a transient occurs. As depicted by the waveform  124 , the interleaved approach for the regulator  40  during the transient produces less voltage drop. Finally, as depicted by the waveform  120 , the best results may be achieved by the non-interleaved approach for the regulator  42  in which the voltage is prevented from decreasing during the transient. 
     Referring to FIG. 16, the regulator  40  may furnish power to a computer system  200 . In this manner, the regulator  40  may be part of voltage regulation circuitry  246  of the computer system  200  and may furnish power to one or more power lines  242 . In addition to the voltage regulation circuitry  246 , the computer system  200  may also include an AC-to-DC converter  240  that may receive an AC wall voltage and convert the AC voltage into a DC voltage that is provided to the voltage regulation circuitry  246 . The voltage regulation circuitry  246  may also receive a DC voltage from a battery pack  243  that furnishes power when AC power is unavailable. 
     Among the components that consume power and may receive the V OUT  voltage, the computer system  200  may include a microprocessor  202  and a bridge circuit, or memory hub  206 , both of which are coupled to a local bus  204 . The memory hub  206  may interface the local bus  204 , a memory bus  209  and an Accelerated Graphics Port (AGP) bus  211  together. The AGP is described in detail in the Accelerated Graphics Port Interface Specification, Revision 1.0, published on Jul. 31, 1996, by Intel Corporation of Santa Clara, Calif. A system memory  208  may be coupled to the memory bus  209 , and a display controller  212  (that controls a display  214 ) may be coupled to the AGP bus  211 . A hub communication link  205  may couple the memory hub  206  to another bridge circuit, or input/output (I/O) hub  210 . 
     The I/O hub  210  includes interfaces to an input/output (I/O) expansion bus  216  and a Peripheral Component Interconnect (PCI) bus  230 . The PCI Specification is available from the PCI Special Interest Group, Portland, Oregon  97214 . An I/O controller  217  may be coupled to the ISA bus  216  and receive input data from a keyboard  224  and a mouse  226 , as examples. The I/O controller  217  may also control operations of a floppy disk drive  222 . A drive controller  231  may be coupled to the PCI bus  230 . The drive controller  231  may control operations of a hard disk drive  232  and a CD-ROM drive  233 , as examples. 
     Other embodiments are within the scope of the following claims. For example, the regulator may be formed from a single Buck converter stage or more than two Buck converter stages in different embodiments of the invention. Furthermore, in other embodiments, a topology (a forward, flyback or a Boost converter topology, as examples) other than a Buck converter topology may be used. A multiple phase converter (three phase or a four phase converter, as examples) other than a two phase converter may be used in other embodiments of the invention. Control schemes, such as a current mode control scheme, may be used other than the voltage mode control scheme that is described above. 
     While the invention has been disclosed with respect to a limited number of embodiments, those skilled in the art, having the benefit of this disclosure, will appreciate numerous modifications and variations therefrom. It is intended that the appended claims cover all such modifications and variations as fall within the true spirit and scope of the invention.