Abstract:
A calibration circuit for calibrating harmonics in a mixer. The calibration circuit includes an RF signal path configured to receive an RF signal corresponding to an output of the mixer, selectively receive a test signal injected into the RF signal path, and provide feedback to the mixer according to the RF signal and/or the test signal. The test signal corresponds to a selected harmonic of the harmonics in the mixer. A measurement circuit is configured to detect, in the output of the mixer, the test signal injected into the RF signal path. A calibration module is configured to receive a feedback signal indicative of the test signal detected in the output of the mixer and, based on the feedback signal indicative of the test signal detected in the output of the mixer, adjust a duty cycle associated with the mixer to calibrate the selected harmonic.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     The present disclosure is a continuation of U.S. patent application Ser. No. 13/774,060 (now U.S. Pat. No. 8,977,211), filed on Feb. 22, 2013, which claims the benefit of U.S. Provisional Application No. 61/604,891, filed on Feb. 29, 2012. The entire disclosures of the applications referenced above are incorporated herein by reference. 
    
    
     FIELD 
     The present disclosure relates to mixers, and more particularly to systems and methods for calibrating harmonic rejection in switching mixers. 
     BACKGROUND 
     The background description provided herein is for the purpose of generally presenting the context of the disclosure. Work of the presently named inventors, to the extent the work is described in this background section, as well as aspects of the description that may not otherwise qualify as prior art at the time of filing, are neither expressly nor impliedly admitted as prior art against the present disclosure. 
     Radio frequency (RF) receivers typically rely on hard-switched mixers to perform down conversion. Down conversion refers to translation of an incoming modulated signal from a higher frequency to a lower frequency. Up conversion from a lower frequency to a higher frequency is also performed when transmitting the signal. 
       FIG. 1  shows an example of a portion of an RF receiver  10  to perform down conversion. The RF receiver  10  includes an antenna  19  that is coupled to an optional RF filter  20  and a low noise amplifier  22 . An output of the amplifier  22  is coupled to a first input of a mixer  24 . A second input of the mixer  24  is connected to a local oscillator (LO)  25 , which provides a reference frequency. The mixer  24  converts radio frequency (RF) signals to intermediate frequency (IF) signals. An output of the mixer  24  is connected to an optional IF filter  26 . The RF receiver  10  may include additional circuits to perform additional down conversion and processing. 
       FIG. 1B  shows an example of a portion of an RF transmitter  50  to perform up conversion. The transmitter  50  includes a variable gain amplifier (VGA)  84  that receives IF signals and that is coupled to an optional IF filter  85 . The optional IF filter  85  is connected to a first input of an IF to RF mixer  86 . A second input of the mixer  86  is connected to an oscillator  87 , which provides a reference frequency. An output of the mixer  86  is coupled to an optional RF filter  88 . The optional RF filter  88  is connected to a power amplifier  89 , which may include a driver. The power amplifier  89  drives an antenna  90  through an optional RF filter  91 . 
     When a frequency range of the incoming modulated signal of the RF receiver has an upper-bound larger than twice a lower bound, harmonic mixing can be an issue. Unwanted signals lying on harmonic frequencies of the local oscillator are down converted and interfere with the wanted signal, which lies on a fundamental frequency of the local oscillator. 
     Harmonic rejection mixers attempt to reduce odd harmonic mixing. Intrinsic device matching is typically exploited to reject even harmonics. However, these approaches are not always sufficient to reduce even and odd harmonics, which adversely impacts silicon yield. 
     As used herein, η refers to duty cycle and η nom  refers to 50% duty cycle. DC offset  refers to DC offset with respect to a predetermined threshold voltage or current value (such as zero). 
     Balanced η refers to duty cycle equal to 50% (or ½) and unbalanced η refers to duty cycle not equal to 50% (or ½). IM n refers to n th  order inter-modulation. As IM n  decreases, linearity increases. IIP n refers to n th  order input referred intercept point. IIP is proportional to 1/IM n . Therefore, linearity increases as IIP n  increases. 
       FIG. 2  illustrates a duty cycle of a square wave signal. The pulse width τ plus  represents a period that the square wave signal is positive while the pulse period T represents a period of the local oscillator square wave. T is related to the fundamental frequency f LO  of the LO as f LO  =1/T. 
     Duty cycle is typically defined as the ratio of the pulse width to the period T and it is quantified as a percentage. For example, if the pulse width to period ratio is ½, then the effective duty cycle is 50%. Under these conditions, the square wave LO spectrum includes only odd harmonics of the fundamental frequency as can be seen in  FIG. 3 . Moreover, these harmonics have energy than decreases with increasing frequency. When the effective duty cycle is not equal to 50%, the LO spectrum also includes even harmonics as can be seen in  FIG. 4 . 
       FIGS. 5-7  illustrate a hard-switching mixer performing down-conversion in a receiver. Incoming RF signal energy at f LO ±f IF  is converted to a lower frequency f IF . The operation produced by the hard-switched mixer can be viewed as a multiplication, in the time domain, between the incoming RF signal and the sign of the local oscillator waveform. The time domain multiplication with the sign(LO) is what differentiates hard-switched mixers from analog multipliers where the RF incoming signal is multiplied with the LO waveform. 
     Because of the periodic and bipolar nature of the LO waveform, sign(LO) is a square-wave with the same period as the original LO waveform. Consequently, if the effective duty cycle of sign(LO) is 50%, the spectrum includes the odd harmonics. If the incoming RF signal includes only energy around the fundamental frequency of the LO (f LO ), the rich harmonic content of sign(LO) is not an issue since only the useful/wanted energy will be translated around f IF  and processed. However if the incoming RF signal has energy around the harmonics of the LO, hard-switched operation may create issues. 
     Referring now to  FIGS. 8-9 , when the incoming RF signal energy spreads close to any of the LO harmonics with some interferers, the nature of the hard-switched mixer downconverts the interferer to IF as well. Harmonic mixing distortion occurs when the interferer energy folds over the useful signal, which corrupts the information in the useful signal. Harmonic mixing distortion is an issue in wireless systems that handle RF bands (incoming for the RX and outgoing for the TX) where the upper bound is 2 or more times the lower bound. 
     When the LO is balanced (or equivalently has a duty cycle of 50%), harmonic distortion should be considered for odd LO harmonics. However, when the LO has a duty cycle that is different than 50%, even order harmonics appear in the sign(LO) spectrum. RF interferers located close to the even LO harmonics fold over the useful signal. Dotted lines in  FIG. 9  illustrates the case where the LO has a duty cycle different than 50%, which leads to the down conversion at f IF  of the interferer placed at f IF +2f LO . 
     SUMMARY 
     A system includes a radio frequency (RF) signal path configured to receive an input signal that includes at least one of a test signal and an RF signal. A local oscillator (LO) signal path is configured to supply a LO signal. A mixer includes a first input in communication with the RF signal path and a second input in communication with the LO signal path. A calibration control module is configured to receive an output of the mixer in response to the test signal and to adjust an effective duty cycle of the LO signal in the LO signal path based on the output of the mixer. 
     In other features, the test signal comprises a tone at one of an odd or even multiple of a fundamental frequency of a wanted signal. The calibration control module adjusts a gain of the RF signal path and a phase of the LO signal path. An analog to digital converter (ADC) module is configured to convert an output of the mixer to a digital signal. A signal processing module is configured to analyze the digital signal and to generate a feedback signal. 
     In other features, the calibration control module receives a feedback signal and adjusts at least one of the effective duty cycle, the gain and the phase based on the feedback signal. The LO signal path includes a DC offset module configured to adjust the effective duty cycle by adjusting a DC offset of the LO signal. The calibration control module communicates with the DC offset module. 
     In other features, the DC offset module includes a first transistor including a first terminal, a second terminal and a control terminal, a second transistor including a first terminal, a second terminal and a control terminal, a first resistor connected between the first terminal of the first transistor and a voltage reference, a second resistor connected between the first terminal of the second transistor and the voltage reference, a first variable current source connected to a node between the first resistor and the first transistor, and a second variable current source connected to a node between the second resistor and the second transistor. 
     In other features, the calibration control module adjusts the gain of the RF signal path, the phase of the LO signal path and the effective duty cycle of the LO signal using a plurality of iterations. The calibration control module adjusts the gain of the RF signal path, the phase of the LO signal path, and the effective duty cycle of the LO signal using a weighting function. 
     Further areas of applicability of the present disclosure will become apparent from the detailed description, the claims and the drawings. The detailed description and specific examples are intended for purposes of illustration only and are not intended to limit the scope of the disclosure. 
    
    
     
       BRIEF DESCRIPTION OF DRAWINGS 
         FIG. 1A  is a functional block diagram of an example of a receiver including a mixer. 
         FIG. 1B  is a functional block diagram of an example of a transmitter including a mixer. 
         FIG. 2  illustrates the effective duty cycle of a square wave. 
         FIG. 3  illustrates odd harmonics in response to a balanced duty cycle. 
         FIG. 4  illustrates even harmonics that occur in response to an imbalanced duty cycle. 
         FIG. 5  illustrates a sign function for a square wave. 
         FIG. 6  illustrates amplitude as a function of frequency for an example RF signal. 
         FIG. 7  illustrates frequency conversion of the signal in  FIG. 6  performed by an ideal hard-switched mixer. 
         FIG. 8  illustrates amplitude as a function of frequency for a wanted signal and an interferer. 
         FIG. 9  illustrates frequency conversion of the signal in  FIG. 8  performed by an ideal hard-switched mixer. 
         FIG. 10  is an electrical schematic of an example of a mixer. 
         FIGS. 11 and 12  illustrate changes in duty cycle due to DC offset. 
         FIG. 13  is a flow chart illustrating an example of a method for calibrating even and odd harmonic rejection according to the present disclosure. 
         FIG. 14  is a flowchart illustrating an example of a method for calibrating even harmonic rejection according to the present disclosure. 
         FIG. 15  is a flowchart illustrating an example of a method for calibrating odd harmonic rejection according to the present disclosure. 
         FIG. 16  is an electrical schematic and block diagram illustrating a portion of a calibration circuit for a mixer according to the present disclosure. 
         FIG. 17  is a functional block diagram illustrating an example of a calibration circuit according to the present disclosure. 
         FIG. 18  is a functional block diagram and electrical schematic of another example of a calibration circuit according to the present disclosure. 
         FIG. 19  is a functional block diagram and electrical schematic of a portion of the calibration circuit of  FIG. 18  for adjusting DC offset. 
     
    
    
     In the drawings, reference numbers may be reused to identify similar and/or identical elements. 
     DESCRIPTION 
       FIG. 10  shows an example of a hard-switched mixer  100 . The mixer  100  includes transistors T 1 , T 2 , T 3  and T 4 . First terminals of transistors T 1  and T 3  are connected together and to a resistor R1. First terminals of transistors T 2  and T 4  are connected together and to a resistor R2. The resistors R1 and R2 are connected to a voltage reference. Second terminals of transistors T 1  and T 3  are connected together and to a first terminal of transistor T 5 . Second terminals of transistors T 2  and T 4  are connected together and to a first terminal of transistor T 6 . Second terminals of the transistors T 5  and T 6  are connected to a current source I. Control terminals of the transistors T 1  and T 2  receive one polarity of the local oscillator signal. Control terminals of the transistors T 3  and T 4  receive the other polarity of the local oscillator signal. Control terminals of the transistors T 5  and T 6  receive the RF signal. 
     Intermodulation is a measure of the mixer linearity performance. When IM 2  rises, mixer linearity decreases. When the LO has a duty cycle that deviates from the optimal 50%, IM 2  and even order intermodulation increases. IIP 2  is a function of non-ideal characteristics or mismatch of the differential structure shown in  FIG. 10 . When the LO square wave does not have a balanced duty cycle and Δη≠0, the imbalance combines with the other mismatch like Δgm, ΔA RF , and ΔR and may lead to a lowering of IIP 2  or equivalently IM 2  rises. In the equation below, α′ 2  is the second-order nonlinearity coefficient relative to an input transconductance stage of the mixer and Δη, Δg m , ΔA RF , and ΔR are the mismatch in percent of the LO switching duty-cycle, transconductance of the mixer driver stage, RF signal amplitude, and load resistance, respectively. In K. Kivekas et al. “Characterization of IIP2 and DC-Offsets in Transconductance Mixer,” IEEE Trans. On Circuits &amp; Systems—II, vol. 48, no. 11, Nov. 2001, it was shown: 
     
       
         
           
             
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     Besides even harmonic distortion, LO duty cycle imbalance has other detrimental effects on the performance of the mixer. If there is noise instead of or in addition to interferers close to the LO harmonics, harmonic mixing may increase noise in the final down converted band at f IF . LO and RF leakage to the output of the mixer may also occur. 
       FIGS. 11 and 12  illustrate duty cycle imbalance is equivalent to DC offset in the original LO(t) waveform for hard-switched mixers. In other words, a DC offset in the original bipolar periodic waveform LO(t) is translated into duty cycle variation in the sign(LO(t)) function. Sign(LO(t)) is the multiplying function of hard-switched mixers. In the examples shown, a positive DC offset in LO(t) translates in a duty cycle &gt;50% for sign(LO(t)). Conversely, a negative DC offset in LO(t) translates in a duty cycle &lt;50% for sign(LO(t)). 
     According to the present disclosure, both odd harmonic and even harmonic calibration are based on similar principles. An injection path in the RF signal path allows insertion of a test signal. Both foreground and background schemes are deployable. For example, for 2nd harmonic calibration, a test signal at 2*f LO  is injected. Different signals are possible, for example pure tones (such as sine signals), pseudo-random signals, etc. can be used. 
     A measurement circuit in the IF path detects the test signal. In some examples, the signal power is measured. Alternately, digital filtering may be applied and then the signal power can be measured. The process is repeated for each harmonic to be calibrated. As can be appreciated, calibration results may affect several harmonics at once. Therefore, an iterative process may be used. 
     Odd harmonic calibration may be achieved by adjusting phase and gain calibration controls, or a subset of them. Search algorithms can be applied to phase and gain controls to minimize a harmonic mixing cost function. When analyzing rejection of the 3rd and 5th harmonics, examples of cost functions include Max(HR3, HR5), HR3 only (HR5 is usually correlated), weighted average of HR3 and HR5, etc. In general, multi-objective optimization may be performed. For example, mixer paths can be calibrated sequentially. Gain and Phase can be calibrated sequentially. Iterations may apply, e.g. calibration of phase, then gain, then phase, etc. 
     Analysis of even order harmonic rejection can be performed by altering the LO duty cycle. In one example, the effective duty cycle can be altered by changing the DC offset of the LO. Calibration can be achieved by injecting a tone at 2*f LO  and minimizing second order harmonic mixing. The power of the down converted tone can be measured. The effective duty cycle can be changed to minimize the measured power. 
     For example, all possible settings of DC offset can be tested. In general, a cost function of even order harmonic mixing can be minimized. Examples of cost functions may include Max(HR2, HR4). In order to achieve concurrent even and odd harmonic calibration, multiple calibration steps may be performed due to interactions between even and odd harmonic calibration. 
       FIGS. 13-15  show examples of methods for harmonic calibration according to the present disclosure. In  FIG. 13 , control sets parameters of the mixer (such as gain, phase and/or DC offset) to default values at  150 . At  154 , even harmonic calibration is performed. At  158 , odd harmonic calibration is performed. At  164 , control determines whether additional iterations are required. If true, control returns to  154 . Otherwise, control ends. As can be appreciated, additional iterations may include either even or odd harmonic calibration rather than both even and odd harmonic calibration. 
     In  FIG. 14 , an example of even harmonic calibration is illustrated. At  170 , the first even harmonic is selected. At  172 , a test signal is injected for the selected even harmonic. At  174 , power is measured in the IF path. At  176 , control determines whether additional even harmonics need to be calibrated. If  176  is true, the next even harmonic is selected at  178  and control returns to  172 . If  176  is false, control uses a weighting function or other mechanism for adjusting DC offset based on the measured power. 
     In  FIG. 15 , an example of odd harmonic calibration is illustrated. At  180 , the first odd harmonic is selected. At  182 , a test signal is injected for the selected odd harmonic. At  184 , power is measured in the IF path. At  186 , control determines whether additional odd harmonics need to be calibrated. If  186  is true, the next odd harmonic is selected at  188  and control returns to  182 . If  186  is false, control uses a weighting function or other mechanism for adjusting gain and phase based on the measured power. 
       FIG. 16  shows an example of a circuit  200  that may be used to cancel harmonic distortion in a mixer. The circuit  200  includes an RF path  202 , which receives a test signal and/or an RF signal. For example only, the test signal may include signals such as tones at a predetermined frequency or more complex test signals. The test signal is input to amplifiers  204 - 1 ,  204 - 2 , . . . , and  204 - n , each including an adjustable gain G 1 , G 2 , . . . , and G n , respectively. Each of the amplifiers  204 - 1 ,  204 - 2 , . . . , and  204 - n  receives one or more branch gain calibration signals to adjust the gains G 1 , G 2 , . . . , and G n . Outputs of the amplifiers  204 - 1 ,  204 - 2 , . . . , and  204 - n  are input to a first input of mixers  208 - 1 ,  208 - 2 , . . . , and  208 - n , respectively. One or more summers  211  may be used to combine outputs of the mixers. 
     A local oscillator path  209  includes a polyphase local oscillator  210  that generates LO 1 , LO 2 , . . . , and LO n , which are input to amplifiers  212 - 1 ,  212 - 2 , . . . , and  212 - n , respectively. Each of the amplifiers  212 - 1 ,  212 - 2 , . . . , and  212 - n  receives a duty cycle and phase calibration signal, which adjusts a duty cycle and/or phase of the local oscillator signal. Outputs of the amplifiers  212 - 1 ,  212 - 2 , . . . , and  212 - n  are input to a second input of mixers  208 - 1 ,  208 - 2 , . . . , and  208 - n , respectively. An intermediate frequency output signal (IF out ) is generated by the outputs of the mixers  208 - 1 ,  208 - 2 , . . . , and  208 - n . 
       FIG. 17  shows an example of a mixer circuit  300  with harmonic rejection that includes the circuit  200  from  FIG. 13  along with additional circuits used to reduce or cancel harmonic distortion. The circuit  300  includes a signal processing module  308  that provides the branch gain calibration to the RF signal path  202  and a control signal to select a phase of the LO. The signal processing module  308  provides control signals to the LO signal path  209  to adjust the polyphase LO. The signal processing module  308  provides feedback to a calibration module  334 . The calibration module  334  provides control signals to a duty cycle and phase regulation module  320 , which adjusts the effective duty cycle and phase of the LO. An output of the mixer  208  is input to an analog to digital converter (ADC)  324 , which converts an analog mixer output to a digital mixer output. The digital mixer output from the ADC  324  is input to the signal processing module  308 . 
       FIG. 18  shows an example of a differential mixer circuit  400  with harmonic distortion reduction. A local oscillator adjustment circuit  402  receives a differential LO signal input, which is amplified by amplifier  404  and output to DC offset module  408 . An output of the DC offset module  408  is input to a mixer  424 . A differential RF signal is input to an amplifier  420 . An output of the amplifier  420  is input to the mixer  424 . A differential output of the mixer  424  is input to a power measuring module  428 , which measures a power level of the differential IF signal. While the power measuring module  428  is shown in a particular location, the power measuring module  428  may be arranged in different locations or performed by other components described herein. An output of the power measuring module  428  is input to an ADC  432 , which converts the differential IF signal to a digital signal. An output of the ADC  432  is input to a signal processing module  436 . 
     The power measuring module  428  provides analog feedback to a calibration module  430 . The signal processing module  436  provides digital feedback to the calibration module  430 . The calibration module  430  provides a control signal to the DC offset module  408  to adjust DC offset of the LO signal. 
       FIG. 19  shows an example of the local oscillator adjustment circuit  402  of  FIG. 18 . The local oscillator adjustment circuit  402  includes transistors T 1  and T 2 . A control terminal of the transistor T 1  receives one input of the differential LO input signal input V in+ and a control terminal of the transistor T 2  receives one input of the differential LO input signal input V in− . First terminals of the transistors T 1  and T 2  are connected by resistors R3 and R4 to a voltage reference. Second terminals of the transistors T 1  and T 2  are connected to a current source  448 . Variable current sources  450  and  452  vary current to first and second nodes  460  and  462  located between the first terminals of the transistors T 1  and T 2  and the respective resistors R3 and R4. The differential local oscillator output is taken at the first and second nodes. Varying the current adjusts the effective duty cycle offset. 
     The foregoing description is merely illustrative in nature and is in no way intended to limit the disclosure, its application, or uses. The broad teachings of the disclosure can be implemented in a variety of forms. Therefore, while this disclosure includes particular examples, the true scope of the disclosure should not be so limited since other modifications will become apparent upon a study of the drawings, the specification, and the following claims. As used herein, the phrase at least one of A, B, and C should be construed to mean a logical (A or B or C), using a non-exclusive logical OR. It should be understood that one or more steps within a method may be executed in different order (or concurrently) without altering the principles of the present disclosure. 
     In this application, including the definitions below, the term module may be replaced with the term circuit. The term module may refer to, be part of, or include an Application Specific Integrated Circuit (ASIC); a digital, analog, or mixed analog/digital discrete circuit; a digital, analog, or mixed analog/digital integrated circuit; a combinational logic circuit; a field programmable gate array (FPGA); a processor (shared, dedicated, or group) that executes code; memory (shared, dedicated, or group) that stores code executed by a processor; other suitable hardware components that provide the described functionality; or a combination of some or all of the above, such as in a system-on-chip. 
     The term code, as used above, may include software, firmware, and/or microcode, and may refer to programs, routines, functions, classes, and/or objects. The term shared processor encompasses a single processor that executes some or all code from multiple modules. The term group processor encompasses a processor that, in combination with additional processors, executes some or all code from one or more modules. The term shared memory encompasses a single memory that stores some or all code from multiple modules. The term group memory encompasses a memory that, in combination with additional memories, stores some or all code from one or more modules. The term memory may be a subset of the term computer-readable medium. The term computer-readable medium does not encompass transitory electrical and electromagnetic signals propagating through a medium, and may therefore be considered tangible and non-transitory. Non-limiting examples of a non-transitory tangible computer readable medium include nonvolatile memory, volatile memory, magnetic storage, and optical storage. 
     The apparatuses and methods described in this application may be partially or fully implemented by one or more computer programs executed by one or more processors. The computer programs include processor-executable instructions that are stored on at least one non-transitory tangible computer readable medium. The computer programs may also include and/or rely on stored data.