Abstract:
The address translation logic of the present invention is incorporated in a global positioning system (GPS) receiver and operates to group data in memory based on translating the address from a direct memory access controller. The data includes post-correlated samples of the correlation of a signal with a generated frequency and a generated code having a plurality of time offsets. In general, the address translation logic organizes the data such that each element of the data associated with particular ones of the plurality of time offsets are grouped together in order to improve the efficiency of performing a fast Fourier transform of the data. In addition, the address translation logic allows the transfer of data from correlation circuitry to memory, from the memory to an FFT module, and from the FFT module to the memory using standard DMA controllers.

Description:
CROSS-REFERENCE TO RELATED PATENT APPLICATIONS  
       [0001]    This U.S. patent application is related to the following concurrently filed U.S. patent applications:  
         [0002]    i) USING FFT ENGINES TO PROCESS DECORRELATED GPS SIGNALS TO ESTABLISH FREQUENCIES OF RECEIVED SIGNALS by Warloe et al.;  
         [0003]    ii) SAVING POWER IN A GPS RECEIVER BY CONTROLLING DOMAIN CLOCKING by Warloe et al.;  
         [0004]    iii) AVOIDING INTERFERENCE TO A GPS RECEIVER FROM WIRELESS TRANSMISSIONS BY TIME MULTIPLEXING GPS RECEPTION by Warloe et al.; and  
         [0005]    iv) IMPROVED GPS RECEIVER by Warloe et al., wherein these related U.S. patent applications are incorporated herein by reference in their entireties. 
     
    
     
       FIELD OF THE INVENTION  
         [0006]    The present invention relates to a GPS receiver, and in particular to address translation logic operating to group post correlation data in memory in order to improve the efficiency of a fast Fourier transform of the data.  
         BACKGROUND OF THE INVENTION  
         [0007]    The global positioning system (GPS) is based on an earth-orbiting constellation of twenty-four satellite vehicles each broadcasting its precise location and ranging information. From any location on or near the earth, a GPS receiver with an unobstructed view of the sky should be able to track at least four satellite vehicles, thereby being able to calculate the receiver&#39;s precise latitude, longitude, and elevation. Each satellite vehicle constantly transmits two signals, generally referred to as L1 and L2. The L1 signal from a satellite vehicle contains a unique pseudo-random noise code ranging signal (C/A code) with a chipping frequency of 1.023 MHz, system data with a bitrate frequency of 50 Hz, and an encrypted precise-code (y-code) with a chipping frequency of 10.23 MHz all being modulated onto a carrier frequency of 1575.42 MHz. The L2 signal consists of the system data and y-code being modulated onto a carrier frequency of 1227.60 MHz.  
           [0008]    In order to calculate a three-dimensional location, a receiver must determine the distance from itself to at least four satellite vehicles. This is accomplished by first determining the location of at least four satellite vehicles using ephemeris data received from the satellites. Once the locations of the satellites have been determined, the distance from the receiver to each of the satellites is calculated based upon the current estimate of receiver position. The measurement of the distance from the receiver to a satellite is based on the amount of time that elapsed between the transmission of a ranging signal from each satellite vehicle and the reception of that chip symbol by the receiver. In particular, the estimated position of the receiver is then corrected based upon a time epoch associated with the received ranging signal.  
           [0009]    In order to acquire the L1 or L2 signal, the receiver must match the C/A code or y-code carried in the L1 signal, or the y-code carried in the L2 signal, with an internally generated code. For the C/A code, this is typically done by correlating the two signals by shifting the generated code through the 1023 possible time offsets of the C/A code until the generated code matches the C/A code carried in the L1 signal. To improve the performance of the search, the generated code may be shifted at shorter intervals than a whole chip. For example, 2046 one-half chip positions may be searched. At the time offset when the generated code matches the C/A code carried in the L1 signal, the two signals will cancel out, leaving only the carrier frequency and system data.  
           [0010]    In addition to finding the time offset of the C/A code or y-code carried in the L1 signal or the y-code carried in the L2 signal, the frequency of the received L1 or L2 signal is typically determined. This may be done by generating a local L1 or L2 signal, and correlating this, together with the generated C/A or Y code with the received signal. Because of the movement of the satellite vehicles relative to the earth, the received frequency will experience a Doppler shift of +/−4,500 Hz from the transmitted frequency of the L1 or L2 signal. Another source of frequency uncertainty is the imperfection of the local oscillator, which typically can add a frequency offset of +/−20 ppm, or +/−30 kHz. However, a good part of this offset is due to variations in temperature, and may be modeled by a GPS receiver with a temperature sensor. With this modeling, the remaining temperature uncertainty could be around 10 kHz. Receiver movement may also cause a Doppler effect, however, this effect is usually insignificant when compared to the movement of the satellite vehicles in a commercial application. Due to the conventional method of the GPS signal detection, the receiver generated L1 or L2 signal needs to be within less that 500 Hz of the received signal for a successful search. Typically the frequency of the generated signal is incremented in 750 Hz intervals as the receiver searches for the correct code/carrier combination.  
           [0011]    Therefore, a two-dimensional search of an approximately 30,000 Hz frequency range and the possible time offsets of the C/A code or the y-code must be made in order to acquire the L1 or L2 signal. Some GPS receivers have been designed to concurrently search all possible time offsets for the C/A code in the L1 signal at a single frequency, thereby requiring an enormous number of correlators. After searching all 1023 or more time offsets at one frequency, the frequency is changed and the process is repeated until a satellite is found or the approximately 30,000 Hz frequency range has been searched. While this approach works well in most cases, new applications for GPS receivers are more likely to have access to a precise time source, narrowing the time, or code position, window that needs to be searched. At the same time, a drive to lower system cost by using cheaper oscillators with larger frequency errors maintains the need to quickly search a wide frequency range. Thus, there remains a need for a GPS receiver capable of concurrently searching the approximately 30,000 Hz range of frequencies to determine the precise frequency of the L1 or L2 signal, while having a modest number of correlators used to determine the time offset of the C/A code or the y-code carried in the L1 or the y-code in carried in the L2 signal.  
         SUMMARY OF THE INVENTION  
         [0012]    The address translation logic of the present invention is incorporated in a global positioning system (GPS) receiver and operates to group data in memory based on translating the address from a direct memory access controller. The data includes post-correlated samples of the correlation of a signal with a generated frequency and a generated code having a plurality of time offsets. In general, the address translation logic organizes the data such that each element of the data associated with particular ones of the plurality of time offsets are grouped together in order to improve the efficiency of performing a fast Fourier transform of the data. In addition, the signal correlated with the generated frequency and the generated code having the plurality of time offsets may be a baseband signal that is a baseband representation of a received GPS signal.  
           [0013]    Those skilled in the art will appreciate the scope of the present invention and realize additional aspects thereof after reading the following detailed description of the preferred embodiments in association with the accompanying drawing figures. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWING FIGURES  
       [0014]    The accompanying drawing figures incorporated in and forming a part of this specification illustrate several aspects of the invention, and together with the description serve to explain the principles of the invention.  
         [0015]    [0015]FIG. 1 illustrates a block diagram of a GPS receiver according to one embodiment of the present invention;  
         [0016]    [0016]FIG. 2 illustrates a block diagram of correlation circuitry associated with a GPS receiver according to one embodiment of the present invention;  
         [0017]    [0017]FIG. 3 illustrates a correlator associated with a GPS receiver according to one embodiment of the present invention;  
         [0018]    [0018]FIG. 4 illustrates data from correlation circuitry during a two-dimensional search for a frequency and time offset of a received signal according to one embodiment of the present invention;  
         [0019]    [0019]FIG. 5 illustrates the functionality of address translation logic associated with a GPS receiver according to one embodiment of the present invention;  
         [0020]    [0020]FIG. 6 illustrates a GPS receiver incorporated in a wireless communications device according to one embodiment of the present invention;  
         [0021]    [0021]FIG. 7 graphically illustrates the output of accumulation circuitry in response to detection of a jamming interference signal according to one embodiment of the present invention; and  
         [0022]    [0022]FIG. 8 illustrates a clock and power management module controlling clock signals associated with exemplary domains of a GPS receiver according to one embodiment of the present invention. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0023]    The embodiments set forth below represent the necessary information to enable those skilled in the art to practice the invention and illustrate the best mode of practicing the invention. Upon reading the following description in light of the accompanying drawing figures, those skilled in the art will understand the concepts of the invention and will recognize applications of these concepts not particularly addressed herein. It should be understood that these concepts and applications fall within the scope of the disclosure and the accompanying claims.  
         [0024]    The present invention is preferably incorporated in a GPS receiver  10 . The basic architecture of a GPS receiver  10  is represented in FIG. 1 and may include a receiver frontend  12 , an antenna  14 , and a digital application specific integrated circuit (ASIC)  16 . The receiver frontend  12  receives information previously modulated on a radio frequency carrier from one or more satellite vehicles through antenna  14 . The received signal is amplified, filtered, downconverted, and digitized by the receiver frontend  12  to produce a digital baseband signal representative of the received signal. The receiver frontend  12  also produces a clock (CLK) signal based on a signal from a local oscillator  17 . The frequency uncertainty of the local oscillator  17  is a major source of the frequency uncertainty of the received signal.  
         [0025]    The digital ASIC  16  processes the digitized baseband signal to extract the information and data bits conveyed in the received signal. Correlation circuitry  18  communicates with a controller  20  to perform such operations as decimation, demodulation, correlation, and accumulation. The controller  20  is interfaced to memory  22 , which may include random-access memory (not shown) and read-only memory (not shown) and may alternatively be internal to the controller  20 . The memory  22  is used by the controller  20  to store GPS related information such as ephemeris data, almanac data, last known position, etc. Further, the memory  22  may store program instructions to be executed by the controller  20 .  
         [0026]    The N parallel outputs from the correlation circuitry  18  are multiplexed by the multiplexer (MUX)  24 , which is controlled by a select signal (SEL) from the controller  20 , into a serial stream of data (DATA) and transferred to addresses in the memory  22 . The addresses where the data is stored are determined by using address translation logic (ATL)  26  to translate addresses from a direct memory access (DMA) controller  28 . Once the data is stored in the memory  22 , fast Fourier transform (FFT) circuitry  30  retrieves the data via the DMA controller  28  and produces transformed data, which is the result of the fast Fourier transform of the data. The result of the FFT process is stored in the memory  22  via the DMA controller  28  for use by the controller  20 . Additionally, the controller  20  is operatively connected to an input/output (I/O) subsystem  32  in order to communicate with external devices.  
         [0027]    Jammer response circuitry  38  provides a control signal (CNTL) to the correlation circuitry  18  when a transmission from a nearby wireless communication device is detected. In another embodiment, the jammer response circuit  38  may be part of a wireless communication device, such as a mobile telephone, capable of asserting the control signal CNTL while transmitting. However, the jammer response circuit  38  may be any circuit or device that is capable of detecting a transmission of a jamming interference signal.  
         [0028]    [0028]FIG. 2 illustrates the correlation circuitry  18  in more detail. The correlation circuitry  18  includes a number of correlators N having been divided into N/4 channels each having four correlators. As an example, a first channel  40  and a last channel  42  each have four correlators  44 ,  46 ,  48  and  50  and  52 ,  54 ,  56  and  58 , respectively. Each of the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  is capable of correlating the baseband signal from the receiver frontend  12  with a generated frequency (F) and a pseudo random noise code having a time offset (OFFSET I ) generated by the controller  20 , where I=0, 1, 2, . . . N−1. Further, each of the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  is controlled by the control signal CNTL from the jammer response circuit  38  such that the correlation process pauses during transmissions from the nearby wireless communication device. While only the first channel  40  and the last channel  42  are illustrated, it should be clear that the correlation circuitry  18  includes N/4 channels, each being essentially the same as the channels  40  and  42  described above.  
         [0029]    A more detailed illustration of each of the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  is given in FIG. 3. Each of the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  may include decimation circuitry  60 , carrier demodulation circuitry  62 , code correlation circuitry  64 , and accumulation circuitry  66 . The decimation circuitry  60  receives the baseband signal from the receiver frontend  12  and decimates a sample rate of the received signal to a decimated rate equal to or less than the sample rate. After decimation, the carrier demodulation circuitry  62  demodulates the decimated baseband signal using the generated frequency F from the controller  20 , thereby providing a demodulated baseband signal to the code correlation circuitry  64 .  
         [0030]    The code correlation circuitry  64  correlates the demodulated baseband signal with the generated pseudo-random noise (PRN) code from the controller  20  having the time offset OFFSET I . Further, each of the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  may demodulate the decimated baseband signal using the same generated frequency F, but may correlate the demodulated baseband signal with the generated code having different time offsets OFFSET I . The output of the code correlation circuitry  64  is accumulated for an amount of time, which depends on the particular design of the GPS receiver  10 , and transferred to the memory  22  via the multiplexer  24 . In one embodiment, the amount of time the output of the code correlation circuitry  64  is accumulated is 32 μs, which is discussed in detail below. The accumulated output of the accumulation circuitry  66  is at a maximum when the frequency F and the time offset OFFSET I  match the frequency and time offset of the baseband signal from the receiver frontend  12 .  
       Establishing the Frequency and Time Offset of GPS Signals  
       [0031]    According to one embodiment, the GPS receiver  10  of the present invention is capable of concurrently searching an approximately 30,000 Hz range of frequencies for the baseband signal received from the receiver frontend  12 . Further, the GPS receiver  10  is capable of performing a two-dimensional search for both the frequency of the baseband signal and the time offset of the C/A code or the y-code carried in the received signal. For this example, the received signal includes up to twelve L1 signals, the baseband signal is a baseband digital representation of the received signal, and the generated code from the controller  20  is the C/A code corresponding to a particular one of the L1 signals. In addition, the number of correlators is 48 (N=48), thereby defining 12 (N/4) channels.  
         [0032]    [0032]FIG. 4 illustrates a data set consisting of the data produced by the correlation circuitry  18  during the two-dimensional search performed by the digital ASIC  16  in the GPS receiver  10 . Each row is the output over time of one of the 48 correlators, examples of which are the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58 . Each column is a partial correlation sample period S 0  . . . S M−1 . Additionally, the data elements DATA X,Y , or partial correlation samples, can be any number of bits, where the subscript X=0, 1, . . . N−1 corresponds to the time offset OFFSET I  and the subscript Y=0, 1, . . . M−1 corresponds to the partial correlation sample periods S 0 , S 1 , . . . S M−1  and M is the number of points in the FFT operation.  
         [0033]    In this example, each of the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  correlate the received signal with the generated frequency F and the generated PRN code having a different time offset OFFSET I  for a total of 2 ms. However, the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  accumulate the results of the correlation and provide the data elements DATA X,Y , also called partial correlation samples, at 32 μs intervals, thereby defining the partial correlation sample periods. By producing 64 partial correlation samples at 32 μs intervals, the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  have effectively correlated the baseband signal with the generated frequency F and the generated PRN code having a different time offset OFFSET I  for a total of 2 ms.  
         [0034]    If each partial correlation sample DATA X,Y  is a 32 μs accumulation of the results of the correlated data, 64 partial correlation samples may be processed by the FFT circuitry  30  by performing a 64-point FFT operation to accomplish a search over an approximately 30,000 Hz frequency range for each of the time offsets corresponding to each of the 48 correlators. The frequency separation, or bin width, of the results of the 64-point FFT operation is 1/(M×T), where M is the number of points in the FFT operation and T is equal to the partial correlation sample period. Therefore, the frequency separation of this 64-point FFT operation is approximately 500 Hz, and the frequency range covered by the operation is approximately 30,000 Hz (64×500 Hz=30,000 Hz). The frequency range covered by the FFT operation corresponds to the approximately 30,000 Hz range of frequencies containing the received signal. Although the two are not centered at the same frequency, the results of the FFT operation can be used to determine the location of the frequency of the received signal within the approximately 30,000 Hz range of frequencies.  
         [0035]    In operation, the two-dimensional search begins when the controller  20  sets the generated frequency F to a nominal frequency associated with the baseband signal from the receiver frontend  12  and sends the generated code with offsets OFFSET 0 , OFFSET 1  . . . OFFSET 47  to the correlation circuitry  18 . It is to be understood that the controller  20  can set the generated frequency F to any of a plurality of frequencies. In addition, the controller  20  is capable of generating a different generated frequency F for each of the channels  40  and  42 .  
         [0036]    Once, the generated frequency F and time offsets OFFSET I  have been sent to the correlation circuitry  18 , the accumulation circuitry  66  of each of the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  accumulates the output of the code correlation circuitry  64  for a the partial correlation period S 0  of the C/A code, thereby producing the partial correlation samples DATA X,0 . In this example, the partial correlation period is approximately 32 μs or 33 C/A code chips. The accumulated outputs of partial correlation samples from the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  are serially transferred by the multiplexer  24  to the addresses in the memory  22  determined by the address translation logic  26 . This process is repeated 64 times for each of the partial correlation sample periods S 0  . . . S M−1  to produce the data set for the 64-point FFT operation performed by the FFT circuitry  30 . A total correlation period for the data set is 2 ms (32 μs×64).  
         [0037]    After the partial correlation samples DATA X,Y  have been stored for each of the partial correlation periods S 0  . . . S M−1  and the offsets OFFSET 0  . . . OFFSET 47 , the data is transferred to the FFT circuitry  30  from the memory  22  using the DMA controller  28 . The FFT circuitry  30  performs the 64-point FFT operation on the data from each of the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  and transfers the results (FFT RESULTS) back to the memory  22  using the DMA controller  28 . This completes one iteration of the two-dimensional search, which has searched the approximately 30,000 Hz range of frequencies and the 48 time offsets. The controller may now determine if the received signal was present at any of the frequency/time/PRN combinations in the data set.  
         [0038]    Several more iterations of the two-dimensional search can be performed to search each possible time offset of the 1023 chip C/A code. For example, if the C/A code is searched in ½ chip steps, 2046 time offsets will be searched. Each iteration searches 48 new time offsets until all time offsets have been searched. After each of the possible time offsets has been searched, the controller  20  can then determine the frequency F and time offset OFFSET I  of the baseband signal from the receiver frontend  12  by processing the results from the FFT circuitry  30  for each iteration. The frequency F and time offset OFFSET I  can be stored in the memory  22  to be accessed by the controller  20 .  
         [0039]    Typically, the GPS receiver  10  will attempt the search for and acquire signals from more than one satellite, each having a different C/A code. Further, the C/A code (or PRN) of the received signals may not be known. Therefore, the GPS receiver  10  may perform more than one successive two-dimensional search. For each successive search, the two-dimensional search described above is repeated with controller  20  sending different generated codes corresponding to possible C/A codes associated with each of the received L1 signals to the correlation circuitry  18 . Once the desired number of two-dimensional searches has been completed, each received L1 signal is then tracked by the GPS receiver  10  using the channels, examples of which are the channels  40  and  42 , where each of the channels is capable of tracking one of the received L1 signals.  
       Address Translation Logic (ATL)  
       [0040]    If the data from only one of the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56 , and  58  were to be transferred to the FFT circuitry  30 , the data transfer could be fully automated with standard DMAs set up by the controller  20 . However, if the data is transferred from the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  in parallel and is multiplexed into the serial stream of data to be transferred to the memory  22  with the DMA controller  28 , the resulting data blocks will have interleaved data from all of the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58 . Without the ATL  26 , the data would need to be re-grouped manually by the controller  20 , increasing the need for system throughput, or de-multiplexed into as many FFT modules as there are correlators. The address translation logic  26  allows the FFT of the data associated with the parallel correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58  to be performed by the single FFT circuitry  30  rather than having numerous of FFT modules processing the data in parallel, or having the controller manually reorganize the data before it is processed by the FFT circuitry  30 . By doing so, the overall size of the GPS receiver  10  and the power consumed by the GPS receiver  10  is reduced.  
         [0041]    The address translation logic  26  translates the addresses from the DMA controller  28  without intervention from the controller  20  such that consecutive data from each of the forty-eight correlators, examples of which are the correlators  44 ,  46 ,  48 ,  50 ,  52 ,  54 ,  56  and  58 , is stored in consecutive memory locations, as illustrated in FIG. 5. By doing so, all of the data relating to a particular time offset OFFSET I  are grouped together in the memory  22 , enabling efficient transfer to the FFT circuitry  30 . For example, the data elements, also referred to as the partial correlation samples, received consecutively from the correlation of the time offset OFFSET 0  are defined as DATA 0,0 , DATA 0,1 , DATA 0,2  . . . DATA 0,M−1 . The address translation logic  26  operates to store these data elements in consecutive locations in the memory  22 . Without the address translation logic  26 , the data from the correlation circuitry  18  would be stored in the order it is received by the memory  22 , which would require the controller  20  to reorganize the data before sending the data to the FFT circuitry  30 .  
         [0042]    Using FIG. 5 as an example, the data elements DATA X,Y  corresponds to the data from the accumulation of the correlation of the received signal with the PRN code having the time offset OFFSET I  and the generated frequency F, where the subscript X corresponds to the time offset OFFSET I  and the subscript Y corresponds to the partial correlation sample period. The data is transferred such that the data is grouped by the partial correlation sample period corresponding to the subscript Y, where Y=0, 1, 2, . . . M−1. For example, the partial correlation samples produced by the correlation of the received signal with the PRN code having each of the time offsets OFFSET I  at the partial correlation sample period S 0 , DATA 0,0 , DATA 1,0 , DATA 2,0 , . . . DATA N−1,0 , are grouped together when received by the memory  22 . Using the translated address from the address translation logic  26 , the memory  22  stores the data transmitted serially from the multiplexer  24  such that the partial correlation samples are grouped by the time offset OFFSET I  corresponding to the subscript X. For example, the partial correlation samples associated with the time offset OFFSET 0  corresponding to the subscript X, DATA 0,0 , DATA 0,1 , DATA 0,2 , . . . DATA 0,M−1 , are grouped together in the memory  22 .  
       Avoiding Interference to a GPS System From Wireless Transmissions  
       [0043]    [0043]FIG. 6 is a simplified block diagram of the GPS receiver  10  being used in combination with a wireless communications device  68 , such as a mobile telephone. The wireless communications device  68  may include receive (RX) circuitry  70 , transmit (TX) circuitry  72 , and control and processing circuitry  74 . The receive circuitry  70  operates to receive the GPS signal and any communication signals. The transmit circuitry  72  operates to transmit communication signals from the wireless communications device  68 . The control and processing circuitry  74  operates to process the communications signals sent to the wireless communications device  68  and send communications data to the transmit circuitry  72  to be transmitted as the communications signals. The receive circuitry  70  and the transmit circuitry  72  are shown to use the antenna  14 , which is also used to receive the GPS signal. However, the receive circuitry  70  and the transmit circuitry  72  may use a separate antenna (not shown) to transmit and receive the communication signals.  
         [0044]    When a jamming signal is strong enough, because of jammer output power and/or close proximity to a GPS receiver  10 , and close enough to the GPS L1 or L2 frequencies, it may pass through the receiver frontend  12  and into the digital ASIC  16  and particularly into the correlation circuitry  18 , where the jamming signal may be tracked as a valid GPS signal. This can cause the tracking loops (not shown) and navigation filters (not shown) of the correlation circuitry  18  and the controller  20  to malfunction, and because these functions incorporate relatively long time constant filters, it may take some time for the GPS receiver  10  to return to normal operation even after the jamming signal is removed.  
         [0045]    The jammer response circuitry  38  detects, or is informed by the control and processing unit  74 , when the transmit circuitry  72  is transmitting the communication signals, which would be a jamming interference signal in the reception of the GPS signal. The communications signals are signals that are transmitted from the wireless communications device  68  under normal operating conditions. Therefore, by using the control signal CNTL from the jammer response circuitry  38 , the digital ASIC has the ability to pause the baseband processing of the very weak L1 or L2 signal, which is typically −133 dBm, while the much stronger communications signal is transmitted from the wireless communications device  68 . The control signal CNTL from the jammer response circuitry  38  allows the accumulation circuitry  66  in the digital ASIC  16  to pause accumulation during a transmission from the transmitter. By doing so, the GPS receiver  10  will only see a minimal performance degradation caused by the transmitted signals from the transmit circuitry  72  of the wireless communications device  68 . The GPS receiver  10  will also return to normal operation much faster once the transmit circuitry  72  of the wireless communications device  68  stops transmitting. This is because the only filters (energy storage elements) that experience the energy from the jamming interference signal are relatively wide bandwidth filters with time-constants of much less than 1 μs (1 C/A chip).  
         [0046]    [0046]FIG. 7 illustrates the effect of the control signal CNTL from the jammer response circuitry  38  on the output of the accumulation circuitry  66 . As illustrated, the accumulation circuitry  66  temporarily stops accumulation when the control signal CNTL is asserted, thereby signifying a transmission of the jamming interference signal. Further, the output of the accumulation circuitry  66  is constant while the control signal CNTL is asserted. When the control signal CNTL signifies the end of the transmission, the accumulation circuitry  66  resumes accumulation. The ability to temporarily stop accumulation during the transmission of a jamming interference signal allows the GPS receiver  10  to maintain system performance while experiencing only a minimal drop in the signal-to-noise ratio.  
       Saving Power by Controlling Domain Clocking  
       [0047]    According to one embodiment, the controller  20  includes a clock and power management (CPM) module  76  as illustrated in FIG. 8. The clock and power management module  76  allows the controller  20  to control the power consumption of the digital ASIC  16  by controlling the clock signals used to clock the digital ASIC  16 . As an example, the digital ASIC  16  can be divided into twelve channel domains, examples of which are a channel 1  domain  78  and a channel 12  domain  80 , an integrated phase modulator (IPM) domain  82 , a data collect domain  84 , an events domain  86 , a user time logic domain  88 , a receiver circuitry domain  90 , and a FFT domain  92  being clocked by clock signals CLK 1  . . . CLK 12 , CLK 13 , CLK 14 , CLK 15 , CLK 16 , CLK 17 , and CLK 18 , respectively. Preferably, each of the domains  78 ,  80 ,  82 ,  84 ,  86 ,  88 ,  90 , and  92  implements complementary metal-oxide-silicon (CMOS) or similar logic such that power consumption ceases when the logic is not clocked.  
         [0048]    The channel domains  78  and  80  include circuitry associated with the channels  40  and  42  and can be powered down when not in use by deactivating the clock signals CLK 1  and CLK 12 , respectively. The IPM domain  82  includes circuitry used by the controller  20  to produce the frequency F and the code having the time OFFSET I  and can be powered down by deactivating the clock signal CLK 13 . The data collect domain  84  includes circuitry for deriving a noise floor used by the controller  20  to determine a relative magnitude of the data from the correlation circuitry  18  with respect to noise received by the receiver  10 , and can be powered down by deactivating the clock signal CLK 14 . The events domain  86  includes logic used to time stamp input or output data received from or sent to the I/O subsystem  32 , and can be powered down by deactivating the clock signal CLK 15 . The user time logic domain  88  includes logic used to keep a local clock (not shown) that is continuously corrected using the received GPS signals, and can be powered down by deactivating the clock signal CLK 16 . The receiver circuitry domain  90  includes circuitry not included in the other domains such as the controller  20 , the address translation logic  26 , and the DMA controller  28 , and can be powered down by deactivating the clock signal CLK 17 . The FFT domain  92  includes the FFT circuitry  30  and can be powered down by deactivating the clock signal CLK 18 .  
         [0049]    The receiver  10  and in particular the digital ASIC  16  of the present invention offer substantial opportunity for variation without departing from the spirit and scope of the invention. For example, the number of correlators N has been shown to be 48 as an example. However, the number N could be any number between 1 and 2046. As another example, the frequency range covered by the 64-point FFT operation is shown to be the approximately 30,000 Hz, but the frequency range could be any range sufficient to overcome errors caused by Doppler and local oscillator imperfections. Further, the number of points in the FFT operation M used to cover the approximately 30,000 Hz range of frequencies could vary depending on particular design requirements. As yet another example, the digital ASIC  16  could be divided into any number of domains, which can be powered down by deactivating the clock signals to the domains.  
         [0050]    The foregoing details should, in all respects, be considered as exemplary rather than as limiting. The present invention allows significant flexibility in terms of implementation and operation. Examples of such variation are discussed in some detail above; however, such examples should not be construed as limiting the range of variations falling within the scope of the present invention. The scope of the present invention is limited only by the claims appended hereto, and all embodiments falling within the meaning and equivalency of those claims are embraced herein.  
         [0051]    Those skilled in the art will recognize improvements and modifications to the preferred embodiments of the present invention. All such improvements and modifications are considered within the scope of the concepts disclosed herein and the claims that follow.