Abstract:
Chopper chopper-stabilized instrumentation and operational amplifiers having ultra low offset. The instrumentation amplifiers use current-feedback, and include, in addition to a main chopper amplifier chain, a chopper stabilized loop for correcting for the offset of the input amplifiers for the input signal and for receiving the feedback of the output voltage sense signal. Additional loops, which may include offset compensation and autozeroing loops, may be added to compensate for offsets in the chopper stabilized loop for correcting for the offset of the input amplifiers. Similar compensation is disclosed for decreasing the offset in operational amplifiers.

Description:
BACKGROUND OF THE INVENTION 
   1. Field of the Invention 
   The present invention relates to the field of instrumentation amplifiers. 
   2. Prior Art 
   An Instrumentation Amplifier is often made up of 3 operational amplifiers (OpAmps). The first two amplifiers are buffer amplifiers. The third amplifier is an amplifier with a four-resistor bridge as a feedback network. This configuration has two main disadvantages: Firstly, the common-mode rejection ratio (CMRR) is limited by the unbalance of the resistive bridge. Secondly, the input voltage common-mode (CM) range cannot include the negative rail because of the overall feedback from the output to the input by the OpAmps (“Operational Amplifiers”, Johan Huijsing, Kluwer Academic Publishers). 
   Therefore, the current-feedback instrumentation amplifier is a better alternative. Its topology is shown in  FIG. 1 . It is excellently suited to allow the negative or positive supply rail voltage to be included into the input common-mode range (“Indirect current feedback instrumentation amplifier with a common-mode input range that includes the negative rail”, B. J. van den Dool et al., IEEE Journal of Solid State Circuits, Vol. 38, No. 7, July 1993, Pgs. 743–749). The reason is that the input signal and feedback signal are independently sensed by the voltage-to-current (V-I) converters G 3  and G 4 . For instance, if these V-I converters are composed of identical differential P-channel pairs, the negative supply rail can be included. For obtaining a better accuracy and CMRR, the V-I converters can be each composed of two high-transconductance composite P-channel transistors with a degeneration resistor between the sources. This also improves the matching of the two identical transconductances G 3  and G 4  for better overall gain accuracy. 
   The instrumentation amplifier of  FIG. 1  further consists of an output stage G 1  and an intermediate stage G 2 . A nested Miller compensation with C M11 , C M12 , C M21 , C M22  provides a preferred straight roll-off of the frequency characteristic. 
   To obtain low offset, choppers can be inserted in the signal path around the input stages, as shown in  FIG. 2 . With choppers, the offset can roughly be reduced by a factor 100–1000, from 10 mV to 100–10 μV. But there are several limitations. Firstly, a square wave at the chopper frequency of the size of the offset referred to the input will appear around the correct average signal value. To erase this square wave, a low-pass filter has to be placed after the instrumentation amplifier. This reduces the bandwidth of the instrumentation amplifier to below 0.1 (10%) of the chopper frequency. If the chopper frequency F 1  is 10 kHz, the bandwidth will be reduced to several hundreds Hz. 
   Secondly, there are several effects that limit the offset reduction. One of them is an imperfect 50% duty cycle of the chopper frequency. Another is an unbalance of the charge injection in the choppers by the switching signal. Further, the initial offset will not fully be averaged out due to parasitic capacitors between the first chopper inputs in combination with attenuation resistors at the inputs. Most of these limitations, except charge injection, would vanish if the initial offset of the input amplifiers could be reduced by trimming or by autozeroing. Trimming is undesirable and not preferred in mass-production due to additional test time, cost and complexity, and lack of stability over temperature and time. One cannot simply autozero an instrumentation amplifier as was done in the prior art for OpAmps (U.S. Pat. No. 6,734,723, Huijsing et al.), because in accordance with  FIGS. 1 and 2 , the input voltage is not zero, but instead, the input stages carry the input and feedback voltages, respectively. In that regard,  FIG. 3  presents a prior art chopper-stabilized OpAmp. Because an OpAmp is a high gain amplifier used with negative feedback, the closed loop differential input voltage to amplifier g 3  is zero, so that the input to chopper Ch 2  is simply the accumulated offsets of amplifiers g 3 , g 2  and g 1  as referred to the input of amplifier g 3 . 
   As used herein and in the disclosure and claims of the present invention to follow, the word stability and the various other forms of the word sometimes refer to stability in the sense of the absence of significant drift over time and temperature, not stability in the sense of absence of self oscillation or ringing, or hangup on either rail. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a block diagram of a prior art current-feedback instrumentation amplifier. 
       FIG. 2  is a block diagram of a prior art instrumentation amplifier like that of  FIG. 1 , though with choppers inserted in the signal path around the input stages. 
       FIG. 3  is a block diagram of a prior art chopper-stabilized OpAmp. 
       FIG. 4  is a block diagram of one embodiment of the present invention chopper chopper-stabilized current-feedback instrumentation amplifier. 
       FIG. 5  is a block diagram of an embodiment similar to that of  FIG. 4 , but including further improvements in the chopper chopper-stabilized current-feedback instrumentation amplifier. 
       FIG. 6  is a block diagram of another embodiment of chopper-stabilized current-feedback instrumentation amplifier. 
       FIG. 7  is a block diagram of an embodiment similar to that of  FIG. 6 , but including further improvements in the chopper chopper-stabilized current-feedback instrumentation amplifier similar to the improvements in the embodiment of  FIG. 5 . 
       FIG. 8  is a block diagram of an improved chopper-stabilized OpAmp. 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   One embodiment of the present invention is shown in  FIG. 4 . The basic chopper current-feedback instrumentation amplifier of  FIG. 2  is used as the main instrumentation amplifier. The voltage-to-current converter G 3  senses the input signal V in =V in+ −V in− , while the voltage-to-current converter G 4  takes the sense feedback output signal V s =V s+ −V s− . If G 3 =G 4 , the high loop gain of the whole amplifier forces the feedback sense voltage V s  to be equal and opposite to the input voltage V in . 
   The choppers Ch 1 , Ch 2  and Ch 3  chop the offset voltage of the amplifiers G 3  and G 4 . The chopped offset can be regarded as a square-wave interference voltage referred to the input voltage of amplifiers G 3  and G 4 . The input voltage V in  is determined by an external source, and while generally may be a varying signal, it does not contain the square-wave signal. The high loop gain of the whole amplifier forces the feedback-sense voltage V s  to compensate the chopped input offset voltage. Therefore, this square-wave chopped input offset will be superimposed on the desired feedback sense voltage V s . 
   In the embodiment of  FIG. 4 , amplifiers G 7  and G 8  (voltage to current converters) are used to obtain a gauge to control the offset of amplifiers G 3  and G 4 . More specifically, with respect to DC levels, the closed loop circuit settles with V in  and V s  being equal and opposite voltages. However the sense voltage V s  has the offset caused square wave on it while V in  does not. Consequently the output current of amplifier G 7  plus the output current of amplifier G 8  will simply be the square wave caused by the offset of amplifier G 3 . Thus the resulting current represents the square-wave chopped input offset voltage component from V s , and largely suppresses the desired input and feedback sense voltages. 
   Next the combined output currents of amplifiers G 7  and G 8  are rectified into a DC current by the chopper Ch 4 . This DC current represents the input offset voltage. Next this DC current is integrated by an integrator amplifier G 6 , with the integrator output voltage being converted into a current by G 5  and added to the output currents of amplifiers G 3  and G 4  in order to gradually cancel the input offset voltage of these amplifiers. Since the offsets are at most very slowly varying, such as by temperature or time variations, in general the response of this offset control loop need not be particularly fast, and generally is intentionally given a time constant much longer than the chopper frequency period so as to be a substantially fixed offset compensation during each chopper period. Note that the integrator has the effect of integrating the rectified square wave on the sense voltage V s , no matter how small, so that, neglecting other sources of error, the offset control loop settles when the offset is eliminated, and is not limited to the gain within the control loop. 
   The chopper chopper-stabilized current-feedback instrumentation amplifier of  FIG. 4  can still be improved on 3 issues. These further improvements are depicted in  FIG. 5 . 
   The sense amplifiers G 7  and G 8  each have an offset voltage. This offset is represented as an offset current at their output and further chopped by chopper Ch 4  ( FIG. 4 ) into a square wave current. This current is integrated into a triangle shaped voltage by the integrator G 6  and added to the output by amplifier G 5 . By chopper Ch 3 , the triangle waveform is reshaped into a sawtooth referred to the feedback sense V s  This is an undesired signal. Also the offset of amplifiers G 7  and G 8  together with an imperfection of the 50% duty cycle of chopper Ch 4  will result in a DC component, which cannot be distinguished from the offset of amplifiers G 3  and G 4 . Therefore, the offset of G 7  and G 8  should be reduced. 
   Thus the first main improvement is to reduce the offset of amplifiers G 7  and G 8 . Therefore, an autozero phase through multiplexer MUX 1  during one full clock cycle is introduced. In this phase the multiplexer allows the output of amplifiers G 7  and G 8  to be integrated by amplifier G 9 . The amplifier G 10  feeds the integrated offset back and corrects for it. 
   The offset of amplifier G 9  should be low because it builds charge across the parasitic capacitances at the output of amplifiers G 7,8 , which will later be discharged by a different offset of the integrator G 6 . This results in an incorrect sensing of the offset of amplifiers G 3  and G 4 , similar to the offset of amplifier G 6 , as described hereinafter, and a square wave residue. To reduce the offset of amplifier G 9 , a chopper stabilisation loop is built around it consisting of the choppers Ch 5  and Ch 6 , the sense amplifier G 11 , the integrator G 12  and correction amplifier G 13 . 
   If integrator G 6  has an input offset voltage, this voltage will show as a square wave before the chopper Ch 4 . This will charge and discharge the parasitic capacitors at the output of amplifiers G 7  and G 8 . These charge pulses will be integrated into a DC voltage by integrator G 6 . This DC voltage cannot be distinguished from the DC integrator voltage that represents the offset of amplifiers G 3  and G 4 . As a result, the offset of amplifiers G 3  and G 4  is not compensated correctly, and a square wave by the choppers Ch 2  and Ch 3  will remain. Therefore, the offset of integrator G 6  has to be reduced. 
   Thus the second main improvement is to reduce the offset of amplifier G 6 . For that purpose, a secondary offset detection and correction circuit has been added similar to the circuitry G 8 , Ch 4 , G 6 , G 5 . The secondary offset sense and correction loop consists of a sense amplifier G 14 , a chopper Ch 7 , an integrator G 15  and a correction amplifier G 16 . The sense amplifier G 14  senses the square wave before chopper Ch 4  caused by the offset of amplifier G 6 . Chopper Ch 7  redirects the square wave and the integrator G 15  integrates the offset caused by amplifier G 6 . The correction amplifier G 16  closes the loop. 
   However, this secondary loop also needs a third order correction. Firstly, the offset of amplifier G 14 , being chopped by Ch 7 , creates a triangle wave at the output of the integrator G 15 . This triangle is added through amplifiers G 16  and G 5  and referred to the feedback input through amplifiers G 3,4  and chopper Ch 3  as a sawtooth waveform. This is undesirable. Therefore, an autozero loop has been placed around amplifier G 14  through multiplexer MUX 2 , integrator G 23  and correction amplifier G 24 . This is similar to MUX 1 , integrator G 9  and correction amplifier G 10 , to correct the offset of amplifiers G 7  and G 8 . 
   The offset of integrator G 15  introduces a square wave before chopper Ch 7 . The parasitic output capacitance at the output of amplifier G 14  creates charge pulses, which are rectified by chopper Ch 7  and integrated again by integrator G 15  into an incorrect correction signal, which looks like an offset of the original integrator G 6 , resulting in a square wave residue. Therefore, another or third order correction loop is created to correct the offset of amplifier G 15 . This loop consists of the sense amplifier G 20 , chopper Ch 8 , integrator G 21 , and correction amplifier G 22 . 
   Finally, the offset of amplifier G 2  in the main amplifier will show as an input offset, but reduced by the voltage gain of amplifiers G 3  and G 4 . If the offset of amplifier G 2  is 10 mV, and the voltage gain of amplifiers G 3  and G 4  is 1000, there still is an offset of 10 μV. Hence it is good to also reduce the offset of amplifier G 2 . 
   Moreover, the offset of amplifier G 2  results in charge peaks introduced by the parasitic capacitances at the output of amplifiers G 3 , G 4  and G 5  in combination with the chopping activity of chopper Ch 1 . Also for this purpose, it is desirable to reduce the offset of amplifier G 2 . 
   The offset of amplifier G 2  results in a residual offset and spikes. Therefore, a sense and correction loop is built around amplifier G 2 , consisting of a sense amplifier G 17 , chopper Ch 9 , integrator G 18 , and correction amplifier G 19 . This is similar as the loop formed by amplifiers G 8,9 , chopper Ch 4 , integrator G 6  and correction amplifier G 5 . 
   It appears possible to simplify the methods hereinbefore described for use in chopper-stabilized amplifiers. A basic architecture for a chopper-stabilized current-feedback instrumentation amplifier is shown in  FIG. 6 . Because there are no choppers in the main feed forward signal path, no square-wave offset related signal can be found at the input voltage VS of amplifier G 4 , though the offsets are still present. 
   However, using choppers Ch 2  and Ch 3  to chop the input voltage V in  and feedback sense voltage V s , and converting the G 8  and subtracting the output currents of amplifiers G 7  and G 8  (V in  and V s  are equal and opposite differential voltages), a current signal representing the chopped offset of amplifiers G 3  and G 4  is obtained. Chopping this again by chopper Ch 4 , a DC signal representing the offset of G 3  and G 4  is obtained. Integrating this signal by integrator G 6  and adding it by a correction amplifier G 5  to the output summing node of amplifiers G 3  and G 4  compensates for the offset. 
   There is one drawback in regard to the chopper chopper-stabilized version of  FIG. 6  however. Specifically, if the gains of amplifiers G 7  and G 8  are not equal, DC input signals at V in  and V s  cannot be distinguished from the offset. Thus the offset correction is DC signal dependent. 
   This can also be interpreted as a gain error ΔA=G 7/8 −G 3/4  at very low frequencies, where the gain of the correction path through G 7  and G 8  and G 6  and G 5  dominates the gain of the straight path through G 3  and G 4 . But these drawbacks may be overcome by auto-trimming or by dynamic-element matching techniques. 
   In the same way as the basic chopper chopper-stabilized instrumentation amplifier of  FIG. 4  was further improved by second-order and third-order correction loops, the chopper stabilized current-feedback instrumentation amplifier of  FIG. 6  can be further if improved. This is shown in  FIG. 7 . Most of the correction loops have been described with respect to  FIG. 5 . The multiplexer MUX 1  together with amplifiers G 9  and G 10  autozero amplifiers G 7  and G 8 , while chopper Ch 5 , amplifier G 11 , chopper Ch 6 , integrator G 12 , and amplifier G 13  chopper stabilize integrator G 9 . Similarly, amplifier G 14 , chopper Ch 7 , integrator G 15  and amplifier G 16  chopper-stabilize integrator G 6 , while multiplexer MUX 2 , integrator G 23  and amplifier G 24  autozero amplifier G 14 , and also amplifier G 20 , chopper Ch 8 , integrator G 21  and amplifier G 22  chopper-stabilize amplifier G 15 . The main purpose of the loop around amplifier G 2  in  FIG. 5  was to reduce the offset of amplifier G 2  so that spikes caused by chopper Ch 1  were reduced. 
   Now that chopper Ch 1  of  FIGS. 3 and 4  has been removed in  FIG. 7 , the chopper-stabilized loop around G 2  might not be necessary anymore. But if in any case this loop is still desired, for instance to reduce the effect of offset of amplifier G 2  on the input, chopper Ch 1  now needs to be placed inside the correction loop together with amplifier G 17 , chopper Ch 9 , integrator G 18  and amplifier G 19 , as shown in  FIG. 7 . 
   The instrumentation amplifier of  FIG. 7  can be reduced to an OpAmp by eliminating amplifier G 4 , chopper Ch 3  and OpAmp of  FIG. 8 . In that regard, the operation of the circuit is identical to that explained with respect to  FIG. 7  with the exception that because it is used as an OpAmp, as explained before, in use, the negative feedback will force the differential input to V in  to zero, so that the only DC component in the input V in  will be the accumulated offsets of amplifiers g 3 , g 2  and g 1  as referred to the input of amplifier g 3 . Consequently cancellation of the DC component of the input signal required in instrumentation amplifiers and accomplished by amplifier G 4 , chopper Ch 3  and amplifier G 8  in  FIG. 7  is not required in the OpAmp of  FIG. 8 . 
   Thus there has been disclosed herein ultra low offset, low spike artifact instrumentation amplifiers that have a main chopper amplifier chain (backwards numbered) amplifiers G 1  and G 2 , chopper Ch 1 , amplifiers G 3,4  and chopper Ch 2,3 , with a first order offset cancellation loop with amplifier G 5 , integrator G 6 , chopper Ch 4  and amplifiers G 7,8 . Also disclosed as possible improvements are up to three second-order cancellation loops comprising; multiplexer MUX 1 , integrator G 9  and amplifier G 10 ; amplifier G 14 , chopper Ch 7 , integrator G 15  and amplifier G 16 ; and amplifier G 17 , chopper Ch 9 , integrator G 18  and amplifier G 19 . Further disclosed as possible improvements are up to three third order cancellation loops; chopper Ch 5 , amplifier G 11 , chopper h 6 , integrator G 12  and G 24 ; and amplifier G 20 , chopper Ch 8 , integrator G 21  and amplifier G 22 . 
   Further disclosed is the application of the inventive aspects of the present invention chopper-stabilized current-feedback instrumentation amplifiers to chopper-stabilized OpAmps. The exemplary embodiments are described with respect to differential amplifiers, though may be realized as single ended amplifiers also, that is, as single input, single output amplifiers. Also in the embodiments disclosed, two output stages are shown, though in some cases, such as in the case of amplifiers that are lightly loaded, a single stage may be used, dispensing with the use of amplifier G 2  and Miller compensation capacitors CM 21  and CM 22 . Also amplifier G 5  may be an attenuator, either an amplifier with a gain of less than one, or simply resistors for converting the integrator output to a current for input to the current summing point or for attenuation. Additional Miller compensated, nested amplifiers may also be incorporated as desired. Thus while certain preferred embodiments of the present invention have been disclosed and described herein for purposes of illustration and not for purposes of limitation, it will be understood by those skilled in the art that various changes in form and detail may be made therein without departing from the spirit and scope of the invention.