Abstract:
A device to regulate current produced by a permanent magnet machine responsive to a plurality of phase current signals. The motor produces torque for application on a shaft. A processing and drive circuit responsive to a direct current command signal and a quadrature current command signal produces phase current signals for input to the motor. A command circuit responsive to the phase current signals, an angular position of said shaft, and a voltage input command signal to produce a direct current error signal and a quadrature current error signal. A control circuit responsive to the direct and quadrature current error signals produces the direct voltage signal command and the quadrature voltage signal command. The control circuit has a direct and quadrature proportional gain, integrator and clamp circuits. An algorithm produces limited or clamped voltage modulation index signals to obtain maximum efficiency and maximum torque per ampere in the speed range. The algorithm ensures that the current regulator does not run out of voltage by limiting the voltage vector to the achievable voltage vector range that provides maximum torque per ampere and maximum efficiency.

Description:
BACKGROUND OF THE INVENTION 
   1. Technical Field 
   The present invention is directed generally to electromechanical machines, and, in particular, to a system and a method for current regulation in the field-weakening operation of permanent magnetic (PM) machines. 
   2. Description of the Related Art 
   In the control of inverter-driven permanent magnet (PM) machines, field-weakening is often used to lower the inverter voltage rating for a given application. That is, without field-weakening the inverter manufacturer would have to use components rated to handle higher levels of voltage for that given application. This would undesirably add incremental costs to the drive system. Field-weakening may be accomplished by configuring the machine windings to provide a greater torque per amp ratio, and thus achieve a lower base speed for a given torque load. During high speed operation, the phase current may be applied to the machine windings in advance of the phase electromotive force. To control the electromotive force, the d-axis current is decreased inversely with speed. 
   Above the base speed of the machine, i.e., where the line-to-line electromotive force voltage due to the magnets has become greater or equal to the source voltage, a field-weakening current is applied to the machine in order to maintain torque. The flux created by this current is in opposition to the rotor flux, and thus reduces the effective electromotive force seen by the inverter. It should be stressed that this current should be carefully regulated to a target value set by the commanded torque and rotor speed. Failure to control the current to an appropriate value will likely result in the application of excessive voltage to the inverter, which is undesirable. 
   It is known to provide a flux weakening algorithm by the use of a number of look-up tables to produce the reference Q-axis and D-axis currents. However, the use of look-up tables requires the creation of numerous and cumbersome data structures within the look-up tables themselves to handle all possible situations in the system and its environment. 
   As further background, as alluded to, operation in the field-weakening region of PM machines may involve commanding the required D-axis current (Ids) as a function of speed. This can be implemented as a look-up (noted above) or as a mathematical function. Another disadvantage, however, of this approach is that if the magnetic characteristics of the PM machine change due to rotor temperature variations and/or the DC-link voltage change, the look-up table or function may no longer be valid. 
   This may cause the current regulator to exceed the voltage limit, which results in the loss of machine control, which is undesirable. 
   Accordingly, there is a need for a control system that minimizes or eliminates one or more of the above-mentioned shortcomings. 
   SUMMARY OF THE INVENTION 
   One object of the present invention is to provide a solution to one or more of the problems as set forth above. The invention relates to system and method for control of permanent magnet (PM) machines. The invention provides additional functionality in the field-weakening region. To extend the operational speed range of PM machines, it is necessary to de-flux the machine by applying a negative Ids current (current in the synchronous D-axis). One advantage of the invention is that it applies the optimal amount of Ids and Iqs at each operating point at and above the base speed of the PM machine. 
   The invention detects when the output of a current regulator exceeds the available voltage vector, and in response thereto, provides for clamping the proportional-integral (PI) current regulators at the available voltage vector. Through the foregoing, the invention ensures that the current regulator does not run out of voltage, maximizes machine efficiency, and provides a maximum possible torque in the field-weakening region. It is important to note that the base speed point may change significantly depending on DC-link voltage variations and on rotor temperature, which can change the magnetic characteristics of the machine. Nonetheless, a controller in accordance with the present invention is able to properly adjust to the variable base speed. 
   A device in accordance with the present invention is provided to regulate current provided to a permanent magnet (PM) machine. The device includes a processing and drive circuit, and a current regulator. The current regulator includes a command circuit, a control circuit, and a limiter. The processing and drive circuit is responsive to a direct voltage command signal (Vds) and a quadrature voltage command signal (Vqs) configured to produce a plurality of phase current signals for input to the PM machine. The command circuit is responsive to a current input command signal. The command circuit is configured to produce a direct current error signal and a quadrature current error signal. The control circuit is responsive to the direct and quadrature current error signals and is configured to produce the direct and quadrature voltage command signals. Finally, in accordance with the present invention, the limiter is configured to limit the direct and quadrature voltage command signals to respective preselected levels. 
   Other features, objects and advantages of the present invention will become apparent to one of ordinary skill in the art from the description that follows and may be realized by means of the instrumentalities and combinations particularly pointed out in the appended claims, taken in conjunction with the accompanying drawings. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The present invention will now be described, by way of example, with reference to the accompanying drawings, in which: 
       FIG. 1  is a schematic and block diagram of a permanent magnet (PM) machine drive employing an exemplary control system according to the invention. 
       FIG. 2  illustrates, in greater detail, an exemplary d-axis current clamp arrangement employed in the circuit of  FIG. 1 . 
       FIG. 3  illustrates, in greater detail, an exemplary q-axis current clamp arrangement employed in the circuit of  FIG. 1 . 
   

   DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   The invention is directed to a method and apparatus for current control in permanent magnet (PM) machines. The method is configured to produce additional functionality in the field-weakening region. 
   To extend the operational speed range of permanent magnet (PM) machines, it is necessary to de-flux the machine by applying additional negative current in the synchronous D-axis, Ids. The advantage of the method is that it applies the appropriate amount of current at each operating point across the entire speed range of the machine, that is, in the constant torque region and in the field-weakening region. 
   An important feature of the invention is that when an output of a current regulator would otherwise exceed the available voltage, the limiter of the invention clamps the voltage vector. 
   It is important to note that the base-speed point of a PM machine may change significantly depending upon variations in rotor temperature and the DC-link voltage. However, the invention is able to properly adjust to the variable base-speed. 
     FIG. 1  shows a block diagram of an exemplary system  10  according to the invention. System  10  includes a device for regulating current in a PM machine  12  and includes a processing and drive circuit  11 , a transform circuit  24  and a current regulator  30 . The device is adapted for controlling a permanent magnet (PM) machine  12 , e.g., a motor/generator, having a stator S and a rotor R for driving a shaft. 
   Processing and drive circuit  11  includes a transform circuit  22 , a space vector modulation block  26 , a pulse-width modulation (PWM) modulator  18 , and an inverter  14 . The machine  12  is driven by a three phrase inverter  14  coupled to a DC-link voltage source  16  (Vdc). Vdc is sometimes hereinafter referred to as the link voltage. A pulse width modulator (PWM)  18  drives the inverter  14  in a known way. 
   Control of the PM machine  12  may be implemented by a digital signal processor (DSP) or the like. Such DSPs are known and are arranged to be responsive to various inputs for producing control outputs, for driving the machine  12  according to the invention. That is, DSPs may be used for implementing one or more of the blocks described herein, in accordance with its respective, described functional requirements. A sensor Sθ is coupled to machine  12  to produce a sensor rotor position (mechanical rotor position) signal θ r . The sensed rotor position signal θ r  is coupled to a conversion block  13  designated “P/2” configured to convert mechanical rotor position into an electrical rotor position θ θ . “P” is the number of machine poles. The electrical rotor position θ θ  is coupled to a pair of coordinate transform circuits  22  and  24 , as shown. The coordinate transform  22  transforms D-axis and Q-axis modulation index signals (V ds  and V qs , sometimes referred to as direct and quadrature voltage command signals) to produce modulation index signals in stationary coordinates α and β. The modulation index signals in the stationary coordinate frame are coupled to and modulated by a space vector modulator  26  in a known manner to produce outputs (designated DUTY a, DUTY b and DUTY c) that drive the voltage PWM modulator  18 . These outputs provide the duty cycle information to PWM modulator  18 . PWM modulator block  18  generates the gate drive signals for inverter  14  for each of the three phases a, b and c, which provides voltage to machine  12 . 
   Drive phase currents signals a, b and c to machine  12  from inverter  14  drive the machine  12 . These are coupled in feedback relation to the coordinate transform  24  which transforms motor drive phase current signals from the inverter  14  to direct and quadrature synchronous feedback axis signals Idsf and Iqsf, respectively. These signals are provided to clamp current regulator  30 . 
   Clamp current regulator  30  is responsive to the direct and quadrature synchronous feedback axis signals Idsf and Iqsf and is configured to produce the direct and quadrature voltage command signals Vds and Vqs. Regulator  30  includes a direct-axis current command block  32 , a quadrature-axis current command block  33 , summers  34  and  36 , and control circuits  38  and  40 . 
   The clamp current regulator  30  includes an Ids current command block  32  and an Iqs current command block  33  which calculate direct and quadrature current command Ids* and Iqs*, respectively (Commands are designated by an asterisk (*)). The current command signals Ids* and Iqs* are summed with the respective direct and quadrature synchronous feedback signals Idsf and Iqsf at summing nodes  34  and  36  respectively. Ids* and Iqs* are each coupled to the non-inverting (+) inputs of respective summing nodes  34  and  36 . Idsf and Iqsf are coupled to the inverting (−) inputs of respective nodes  34  and  36 . The summed signals output from summers  34 ,  36  respectively represent the D-axis current error signal Id error  or and the Q-axis current error signal Iq error . The error signals are coupled to corresponding proportional-integral (PI) control circuits  38  and  40 . The outputs of the PI control circuits represent the D-axis voltage Vds and the Q-axis voltage Vqs respectively. These signals are coupled to the transform circuit  22  for appropriate transformations as noted above. The voltage signals Vds and Vqs are also fed back to the Ids current command block  32  as shown. 
   The proportional-integral control circuits  38  and  40  (i.e., PI current regulators) include limiters for clamping the D-axis and Q-axis modulation index signals Vds and Vqs to predetermined voltage levels to prevent an undesirable loss of current regulation, to maximize machine efficiency and to provide maximum torque in the field-weakening region. 
   The current command circuit  32  receives a Torque* command input, and a zero vector time control command signal T 0 * as an input. Alternatively, the current command circuit  32  may receive, in lieu of T 0 *, a quantity corresponding to the addition of a time 1 vector (T 1 ) and a time 2 vector (T 2 ) as a control signal, i.e., designated as (T 1 +T 2 )* in the drawings. In a still further embodiment, again in lieu of T 0 * or (T 1 +T 2 )*, circuit  32  may receive a voltage magnitude command signal, designated V MAG *. Note that T 0 +T 1 +T 2 =1. The Ids current command circuit  32  receives a feedback signal corresponding to the zero vector time T 0  from space vector modulator  26 . The feedback signal from space modulator  26  may alternatively be in the form of the sum of vectors  1  and  2  i.e., (T 1 +T 2 ) or according to a known relationship, namely V MAG =√{square root over (Vds 2 +Vqs 2 )}. 
     FIG. 2  illustrates in greater detail the Ids current command circuit  32  and control circuit  38 . First, circuit  32  will be described in greater detail. As illustrated, for the D-axis, the zero time vector T 0  is fed back from space vector modulator  26  and is input to a ripple filter  42 . The output of filter  42  is a time zero feedback output T 0   f  which is provided to the non-inverting (+) input of summer  44 . The zero time vector control command signal T 0 * is coupled to the inverting input (−) of the summer  44 . The difference produced by and output from summer  44  is an error signal T 0   e  which is coupled to a proportional integrator (PI) circuit  46 , whose output feeds limiter or clamp  48 . The clamp  48  has a feedback loop of  50 , which is coupled to the PI circuit  46  as shown. The limiter  48  produces an output designated IdsT 0 , which is limited to values less than or equal to zero in a preferred embodiment, although more generally, such limiter may limit between−Variable (lower)≦. IdsT 0 ≦Variable (upper). 
   The output IdsT 0  of limiter  48  is summed at node  54  with a look up table (IdsLUT) output for Ids from a maximum torque per ampere curve block  52 , both of which are non-inverted, as shown, to produce the D-axis current command control signal Ids*. Block  52  is configured to receive the TORQUE* command control signal. 
   Turning now to the circuit  38  in  FIG. 2 , the Ids* signal and Idsf feedback signal are summed at the non-inverting (+) and inverting (−) inputs of node  34 , respectively, to produce an error signal designated Id error, which is coupled to parallel connected proportional gain circuit  56  and an integrating function circuit  58 . Proportional circuit  56  produces an output that is coupled to the non-inverting input (+) of summing node  60 . The integrating circuit  58  includes an input coupled to node  34 . Circuit  58  also includes an input configured to receive a feedback signal from a clamp or limiting circuit  62  having a feedback loop  64  as shown. The output of circuit  58  is provided to clamp or limiter  62 . 
   The proportional gain circuit  56  controls, among other things, the transient components of the Id error signal, and the integrating circuit  58  controls, among other things, steady state components of the Id error signal. The clamp  62 , when implemented, is used to limit the steady state value within an allowable range (e.g., −Vmag*≦Vds≦Vmag*). The output of the clamp  62  is coupled to another non-inverting input (+) of summing node  60 . The output of node  60  is the unclamped D-axis modulation index signal Vds. This signal is coupled to clamp or limiting circuit  66 , and when engaged, the output of the clamp  66  is the clamped D-axis modulation index signal Vds. This may correspond as well to the range: −Vmag*≦Vds≦Vmag*. As shown in  FIG. 1 , the Vds signal is coupled to the transform circuit  22  and is fed back to the current command circuit  32 . The clamp  66  limits the overall output Vds. 
     FIG. 3  shows, in greater detail, Q-axis control circuit  40  (i.e., PI current regulator). Iqs current command block  33  produces an output, thereof the Iqs* command control signal, in response to the torque command signal TORQUE* and feedback signal Idsf from transform block  24 . The Q-axis reference or command signal Iqs* is coupled to the non-inverting input (+) of the node  36 . The Iqsf feedback signal from transform circuit  24  is coupled to the inverting input (−) of the node  36 . The node  36  produces an error signal designated Iq error  signal. The Iq error signal is coupled to a proportional gain controller  70  and an integrator function block  72 . Proportional gain circuit  70  controls, among other things, transient signals and the integrating circuit  72  controls, among other things, steady state signals. The output of the proportional gain circuit  70  is coupled to the non-inverting input (+) of a summing node  74 . The output of the integrator  72  is coupled to a clamp or limiter  76 . The clamp  76 , when implemented, is used to limit the steady state value within an allowable range defined as follows:
 MOTORING MODE: MIN≦ Vqs ≦└√{square root over ( V mag *2   −Vds   2 )}┘* K   GENERATING MODE: MIN≦ Vqs≦V mag* K           where MIN is a lower limit, and which is preferably zero.       
   The output of the clamp  76  is coupled to another non-inverting input (+) of summing node  74  and is fed back over feedback loop  78  to integrator  72 , as shown. The signals are summed at node  74 , and the output of the node  74  is the unclamped Vqs. This signal is coupled to a clamp or limiting circuit  80 , and when engaged, the output of the clamp is a clamped Q-axis modulation index signal Vqs. This, in turn, is coupled to the transform circuit  22  and is fed back to the current command circuit  32  as shown in  FIG. 1 . The clamp  80  limits the overall output Vqs in accordance with the following equations:
 
MOTORING MODE: MIN≦ Vqs ≦└√{square root over ( V mag *2   −Vds   2 )}┘* K 
 
GENERATING MODE: MIN≦ Vqs ≦( V mag*) K 
 
These equations limit Vqs in the motoring mode to a quadrature voltage command signal limit value derived in the motoring mode from the square root of the quantity ((Vmag*) 2 −(Vds) 2 ) and derived in the generating mode from the voltage magnitude command signal Vmag*.
 
   When activated, clamps  66  and  80  limit the overall D and Q axis signals Vds and Vqs. Clamps  62  and  76  limit or clamp the steady state signals components of the Vds and Vqs signals. 
   Alternatively, Vds and Vqs limits can be calculated as follows: 
   If Delta is greater than Delta Maximum, Vds and Vqs are recalculated as follows:
 
 Vds=−V mag*[ sin(Delta Maximum)],
 
 Vqs=V mag*[ cos(Delta Maximum)],
 
   If Delta is less than Delta Minimum, Vds and Vqs are recalculated as follows:
 
 Vds=−V mag*[ sin(Delta Minimum)],
 
 Vqs=V mag*[ cos(Delta Minimum)],
 
   Where Delta is defined as follows:
 
Delta=arctan(− Vds/Vqs )
         Delta must be within the following range:
 
Delta Minimum≦Delta≦Delta Maximum.
       

   Activation of clamps  66  and  80  occurs when the unclamped Vds and unclamped Vqs are out of their respective, selected voltage vector ranges, i.e., magnitude and direction (delta angle). Once engaged, clamps  66  and  80 , with corresponding clamps  62  and  76 , are operative through feedback lines  67  and  81  to implement an algorithm according to the invention to limit Vds and Vqs to the clamped values shown. 
   From the foregoing, it can be seen that a new and improved device to regulate current consumed by a PM machine has been provided. It is to be understood that the description of the exemplary embodiments is merely illustrative of some of the many specific embodiments that represent applications of the principles of the present invention. Other arrangements would be evident to those skilled in the art without departing from the scope of the invention as defined by the following claims.