Abstract:
A system and method are provided for direct-conversion of a modulated radio-frequency (RF) signal. After receiving an RF signal, the RF signal is mixed with a plurality of oscillator signals with different phases in an interleaving manner.

Description:
RELATED APPLICATION(S) 
   This application claims the benefit of U.S. Provisional Application No. 60/456,510, filed Mar. 24, 2003, entitled DIRECT CONVERSION RECEIVER WITH QUADRATURE BALANCING AND DC OFFSET REMOVAL. 

   FIELD OF THE INVENTION 
   The present invention relates to circuitry, and more particularly to direct-conversion circuitry. 
   BACKGROUND OF THE INVENTION 
   Direct-conversion is a wireless receiver architecture particularly suited to highly integrated, low-power terminals. Its advantage over traditional superheterodyne architectures is that the received signal is amplified and filtered at baseband rather than at some higher intermediate frequency. This architecture results in lower current consumption in the baseband circuitry and a simpler frequency plan. 
   In direct-conversion receivers, the most serious drawback is that the direct current (DC) offset generated by the down-conversion mixers and baseband circuitry. This offset appears in the middle of the down converted signal spectrum, corrupting the signal. 
   The first cause of DC offset is the transistor mismatch of the baseband components such as the down-conversion mixers and buffers. This is static DC offset. In addition, there is dynamic DC offset. One source of dynamic DC offset occurs when the local oscillator (LO) leaks into the front end of the receiver through the integrated circuit substrate. This signal is down converted to DC. Another source of dynamic DC offset occurs when the LO leaks out the antenna and reflects off external objects and back into the receiver. This too is down converted to DC. 
   DC offsets may be removed through capacitive coupling if the signal modulation is tolerant to the phase distortion cause by capacitor-resistor (CR) coupling. In addition, DC offsets may be estimated and digitally removed at the cost of additional hardware size and complexity. 
   Another problem in direct-conversion receivers is in-phase and quadrature (“IQ”) imbalance of the LO and receiver. In the art, it is well known that direct-conversion transmitter and receivers need a local oscillator with quadrature outputs for vector modulation and demodulation. However, when the quadrature outputs are not equal in amplitude and not exactly 90 degrees out of phase, demodulation becomes more difficult requiring a higher signal-to-noise ratio to properly decode the signal. 
   Quadrature phases are typically derived by passing a reference local oscillator through a CR-RC phase shift network. Ideally, this creates two signals with equal amplitude and 90 degrees of phase difference. However, this depends on the accuracy of resistors and capacitors which make up the phase shift network. The resistors and capacitors can vary by up to 15 percent in a typical integrated circuit causing the in-phase and quadrature components to have different amplitudes and a phase difference not equal to 90 degrees. 
   In addition, layout differences between the in-phase and quadrature paths can cause additional amplitude/phase imbalance. Contributing to further in-phase/quadrature imbalance is the circuits in the in-phase and quadrature paths, such as amplifiers and mixers, the physical properties of which differ slightly. Many feedback calibration schemes have been proposed and implemented to mitigate quadrature imbalance at the cost of hardware and/or system complexity. 
   In addition to DC offset and quadrature imbalance, radio-frequency (RF) integrated circuits suffer from self-generated interference. Specifically, signals from one part of the integrated circuit couple to another part of the integrated circuit. The RF section of an integrated circuit is the most susceptible portion since the received signal has not been fully amplified. One way to combat this problem is to turn the signal from single-ended to differential. A differential signal is comprised of a negative and a positive component. This adds to the signal&#39;s resilience to self interference. 
   A conventional direct-conversion receiver is illustrated in  FIG. 1 . As illustrated in  FIG. 1 , a direct-conversion receiver takes an RF signal  10  characterized by a modulation bandwidth and a center frequency. The LO produces a sinusoidal signal which has the same frequency as the RF signal center frequency, as is typical for direct-conversion receivers. As an example, a Bluetooth™ signal might be transmitted at 2440 MHz therefore the LO may produce a 2440 MHz sinusoidal signal for down conversion. 
   Furthermore, the receiver multiplies the RF signal not with one but with two different phases  11 ,  12  of the LO. The two phases  11 ,  12  of the local oscillator are 90 degrees apart and thus, are known as the in-phase (I)  11  and quadrature (Q)  12  components. Through this disclosure, the in-phase local oscillator signal is denoted LO I  and the quadrature local oscillator signal is denoted LO Q . The mixer outputs  13 ,  14  are known as baseband signals since they are at a lower frequency than the RF signal. The baseband signals are in-phase and quadrature corresponding to the in-phase and quadrature local oscillator signals. The baseband signals are low pass filtered as to remove unwanted interfering signals. Through this disclosure, the in-phase baseband signal is denoted BB I  and the quadrature baseband signal is denoted BB Q . The resulting filtered baseband signals  15 ,  16  can be represented by Equations 1 and 2.
 
 BB   I   =RF×LO   I   Equation 1
 
 BB   Q   =RF×LO   Q   Equation 2
 
   Another conventional direct-conversion architecture is shown in  FIG. 2 . This differential direct-conversion architecture is more resilient to self-generated noise than the one illustrated in  FIG. 1 . In  FIG. 2 , the RF input signal  200  is converted by a balun  220  to a differential signal composed of positive and negative components  201 ,  202  respectively. The relationship between the RF input  200  and the differential components  201 ,  202  are described by Equation 3.
 
 RF =( RF   pos   −RF   neg )  Equation 3
 
   Similarly, the differential direct-conversion architecture shown in  FIG. 2  uses differential LO signals to mix the RF signal down to baseband. The polyphase network  205  is a circuit which converts the local oscillator&#39;s voltage waveform  203  into four voltage waveforms  206 ,  207 ,  208 ,  209  at the same frequency as the LO  203  but at 0, 180, 90, 270 degrees offset compared to the LO signal  203  respectively. 
   Collectively, these four signals  206 ,  207 ,  208 ,  209  are referred to as polyphase local oscillator signals. To facilitate the description of this embodiment, these signals are denoted  206 ,  207 ,  208 ,  209  as LO 0 , LO 180 , LO 90 , LO 270  corresponding to their phase shift compared to the local oscillator  203 . It is well known in the art that shifting a sinusoidal signal 180 degrees in phase is the same as inverting the signal. Therefore, the equivalent single-ended in-phase and quadrature LO signals are described mathematically as in Equations 4 and 5.
 
 LO   I   =LO   0   −LO   180   Equation 4
 
 LO   Q   =LO   90   −LO   270   Equation 5
 
   The differential RF signal  201 ,  202  is then routed to the differential mixers  210 ,  211  where it is multiplied by the differential local oscillator signals. At the first mixer  210 , the differential RF signal is multiplied by the in-phase LO (LO I ) to generate the differential in-phase baseband signal  212 ,  213  (BB I ). Likewise, at the second mixer  211 , the differential RF signal is multiplied by the quadrature LO (LO Q ) to generate the differential quadrature baseband signal  214 ,  215  (BB Q ). Equations 6 and 7 describe the mixing process of the differential signals to generate the BB I  and the BB Q .
 
 BB   I =( BB   I,pos   −BB   I,neg )=( RF   pos   −RF   neg )×( LO   0   −LO   180 )  Equation 6
 
 BB   Q =( BB   Q,pos   −BB   Q,neg )=( RF   pos   −RF   neg )×( LO   90   −LO   270 )  Equation 7
 
   As in the single-ended case, the baseband signals  212 ,  213 ,  214 ,  215  can be filtered to remove unwanted interfering signals to produce filtered baseband signals  216 ,  217 ,  218 ,  219 . 
   Now, to elucidate the problems with direct-conversion receivers, DC offset and imbalance distortions will be added to Equations 6 and 7. DC offsets are added to the output of the mixers. DC  1  represents the differential DC offset of the first mixer  210  and DC 2  represents the differential DC offset of the second mixer  211 . Likewise the amplitude and phase imbalance of the mixers and the polyphase LO signals can be accounted for at the output of each mixer. A complex multiplicative term, A 1 e jP1 , represents a random amplitude variation (A 1 ) and a random phase variation (P 1 ) introduced by the first mixer  210  and the signal path and LO path connected to the mixer. Likewise, A 2 e P2  represents a random amplitude and phase variation introduced by the second mixer  211  and the signal and LO paths connected thereto. Thus, with these distortions added, Equations 6 and 7 become Equations 11 and 12.
 
 BB   I =( RF   pos   −RF   neg )×( LO   0   −LO   180 )× A 1 e   jP1   +DC 1  Equation 11
 
 BB   Q =( RF   pos   −RF   neg )×( LO   90   −LO   270 )× A 2 e   jP2   +DC 2  Equation 12
 
   As seen in Equations 11 and 12, the baseband in-phase and quadrature signals imbalance grows as A 1  and A 2  differ and as P 1  and P 2  differ. As the imbalance increases, it is harder for the signal to be received and decoded. Likewise, as DC 1  and DC 2  get larger, and thus depart from the ideal of no DC offset, it becomes more difficult for the signal to be received and decoded. 
   SUMMARY OF THE INVENTION 
   A system and method are provided for direct-conversion of a modulated radio-frequency (RF) signal. After receiving an RF signal, the RF signal is mixed with a plurality of oscillator signals with different phases in an interleaving manner. 
   In one embodiment, the RF signal may be converted to a differential RF signal. Further, the RF signal may be modulated over a finite bandwidth. 
   In another embodiment, the oscillator signals may include an oscillator signal frequency substantially equal to an RF signal frequency of the RF signal. Optionally, the oscillator signals may have phase differences of 0, 90, 180 and 270 degrees. 
   In still another embodiment, the mixing may be carried out by a plurality of mixers. Further, the oscillator signals may be input to the mixers in the interleaving manner. For example, the oscillator signals may be input to the mixers in the interleaving manner by switching which oscillator signals are input to which mixers. 
   As an option, such switching may occur at a rate that is faster than a bandwidth of the RF signal. Further, the switching may occur in a substantially random manner, or even in a completely random manner. 
   In still yet another embodiment, a modulation of the RF signal may be reconstructed as a quadrature baseband signal and an in-phase baseband signal with a de-interleaving operation. Optionally, such de-interleaving operation may include inverting and routing operations. 
   Still yet, low-pass filtering may be applied to the in-phase baseband signal and the quadrature baseband signal. 
   In use, a direct current (DC) offset of the in-phase baseband signal and the quadrature baseband signal may thus be removed. Further, an amplitude and a phase distortion in the in-phase baseband signal and the quadrature baseband signal may be equated or reduced. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a block diagram of a conventional single-ended direct-conversion receiver, in accordance with the prior art. 
       FIG. 2  is a block diagram of a conventional differential direct-conversion receiver, in accordance with the prior art. 
       FIG. 3  is a block diagram of a differential direct-conversion receiver with local-oscillator phase interleaving and baseband de-interleaving, in accordance with one embodiment. 
       FIGS. 4   a–d  show the four configurations of the local-oscillator phase interleaver of  FIG.3 . 
       FIGS. 5   a–d  show the four configurations of the baseband de-interleaver of  FIG. 3 . 
       FIG. 6  is an illustration of a wireless communication system in which one embodiment may be used. 
       FIG. 7  illustrates the functional components of a wireless communication device, shown in block diagram format. 
   

   DETAILED DESCRIPTION 
   Turning to one embodiment in  FIG. 3 , a differential direct-conversion architecture and associated method are provided with two switching matrices:  1 ) the local oscillator phase interleaver (LOPI)  310  circuit, and  2 ) the baseband de-interleaver (BBDI)  330 . Each has four combinations of connections. The four combinations of LOPI are illustrated in  FIGS. 4   a ,  4   b ,  4   c  and  4   d . In keeping with the terminology established in the previous example, the polyphase LO signals  306 ,  307 ,  308 ,  309  are denoted: LO 0 , LO 90 , LO 180 , LO 270 . 
   To establish additional notation to unify the description the signals in  FIG. 3  and  FIG. 4 , in  FIG. 3  the positive input  313  to the first mixer  341  will be denote M 1 + in  FIG. 4 . Likewise, in  FIG. 3  the negative input  314  to the first mixer  341  will be denoted M 1 − in  FIG. 4 . Likewise, in  FIG. 3  the positive input  315  to the second mixer  340  will be denoted M 2 + in  FIG. 4 . Likewise, in  FIG. 3  the negative input  316  to the second mixer  340  will be denoted M 2 − in  FIG. 4 . 
   In state  1  as illustrated in  FIG. 4   a , the local oscillator phase interleaver passes LO 0 , LO 180 , LO 90 , LO 270  through to M 1 +, M 1 −, M 2 +, M 2 − respectively. In the second state as illustrated in  FIG. 4   b , the LO phase interleaver routes LO 180 , LO 0 , LO 270 , LO 90  through to M 1 +, M 1 −, M 2 +, M 2 − respectively. In state  3  as illustrated in  FIG. 4   c , LO 90 , LO 270 , LO 0 , LO 180  are routed to M 1 +, M 1 −, M 2 +, M 2 − respectively, Finally in state  4  as illustrated in  FIG. 4   d , LO 270 , LO 90 , LO 180 , LO 0  are routed to M 1 +, M 1 −, M 2 +, M 2 − respectively. 
   Through this method, each mixer input sees each polyphase LO signal LO 0 , LO 90 , LO 180 , LO 270 . Thus, if a phase or amplitude imbalance of one mixer distorts one LO component then it distorts all components. For the four states, the output of the first mixer  341  is described by Equations 13, 14, 15 and 16, and the output of the second mixer  340  is described by Equations 17,18, 19, and 20.
 
State 1: ( RF   pos   −RF   neg )×( LO   0   −LO   180 )× A 1 e   jP1   +DC 1  Equation 13
 
State 2: ( RF   pos   −RF   neg )×( LO   180   −LO   0 )× A 1 e   jP1   +DC 1  Equation 14
 
State 3: ( RF   pos   −RF   neg )×( LO   90   −LO   270 )× A 1 e   jP1   +DC 1  Equation 15
 
State 4: ( RF   pos   −RF   neg )×( LO   270   −LO   90 )× A 1 e   jP1   +DC 1  Equation 16
 
State 1: ( RF   pos   −RF   neg )×( LO   90   −LO   270 )× A 2 e   jP2   +DC 2  Equation 17
 
State 2: ( RF   pos   −RF   neg )×( LO   270   −LO   90 )× A 2 e   jP2   +DC 2  Equation 18
 
State 3: ( RF   pos   −RF   neg )×( LO   0   −LO   180 )× A 2 e   jP2   +DC 2  Equation 19
 
State 4: ( RF   pos   −RF   neg )×( LO   180   −LO   0 )× A 2 e   jP2   +DC 2  Equation 20
 
   In the context of the present description, “interleaving” may refer to the plain and ordinary meaning thereof, as well as any sort of switching, exchanging, toggling, swapping, interchanging, etc. 
   The BBDI  330  undoes the interleaving that the LOPI introduced. The baseband de-interleaver  330  circuit interleaves between one of four combinations of connections illustrated in  FIGS. 5   a–d . In addition, in two of the states it inverts the incoming signal. To clarify the corresponding notations between  FIG. 3  and  FIG. 5 , in  FIG. 3  the positive output  317  of the first mixer  341  corresponds to the notation B 1  in  FIG. 5 . Likewise, in  FIG. 3  the negative output  318  of the first mixer  341  corresponds to the notation B 2  in  FIG. 5 . Likewise, in  FIG. 3  the positive output  319  of the second mixer  340  corresponds to the notation B 3  in  FIG. 5 . Likewise, in  FIG. 3  the negative output  320  of the second mixer  340  corresponds to the notation B 4  in  FIG. 5 . 
   To further clarify the corresponding notations between  FIG. 3  and  FIG. 5 , in  FIG. 3  the positive in-phase baseband input  332  to the first low pass filter  342  corresponds to the notation B 5  in  FIG. 5 . Likewise, in  FIG. 3  negative in-phase baseband input  333  to the first low pass filter  342  corresponds to the notation B 6  in  FIG. 5 . Likewise, in  FIG. 3  positive quadrature baseband  334  input to the second low pass filter  343  corresponds to the notation B 7  in  FIG. 5 . Likewise, in  FIG. 3  negative quadrature baseband input to the second low pass filter  342  corresponds to the notation B 8  in  FIG. 5 . 
   In state  1  illustrated in  FIG. 5   a , the baseband de-interleaver passes B 1 , B 2 , B 3 , B 4  through to B 5 , B 6 , B 7 , B 8  respectively. In the second state illustrated in  FIG. 5   b , the baseband de-interleaver inverts the incoming signals and routes B 1 , B 2 , B 3 , B 4  to B 5 , B 6 , B 7 , B 8  respectively. In state  3  illustrated in  FIG. 5   c , B 3 , B 4 , B 1 , B 2  are routed to B 5 , B 6 , B 7 , B 8  respectively. Finally, in state  4  illustrated in  FIG. 5   d , the baseband de-interleaver inverts the incoming signals and routes B 3 , B 4 , B 1 , B 2  to B 5 , B 6 , B 7 , B 8  respectively. 
   Returning to  FIG. 3 , the in-phase baseband signal is comprised of differential signals  332 ,  333 . Likewise the quadrature baseband signal is comprised of differential signals  334 ,  335 . For the four states, the in-phase baseband signal is described by Equations 21, 22, 23, and 24. Likewise, the quadrature baseband signal is described by Equations 25, 26, 27, and 28.
 
State 1: ( RF   pos   −RF   neg )×( LO   0   −LO   180 )× A 1 e   jP1   +DC 1  Equation 21
 
State 2: ( RF   pos   −RF   neg )×( LO   0   −LO   180 )× A 1 e   jP1   −DC 1  Equation 22
 
State 3: ( RF   pos   −RF   neg )×( LO   0   −LO   180 )× A 2 e   jP2   +DC 2  Equation 23
 
State 4: ( RF   pos   −RF   neg )×( LO   0   −LO   180 )× A 2 e   jP2   −DC 2  Equation 24
 
State 1: ( RF   pos   −RF   neg )×( LO   90   −LO   270 )× A 2 e   jP2   +DC 2  Equation 25
 
State 2: ( RF   pos   −RF   neg )×( LO   90   −LO   270 )× A 2 e   jP2   −DC 2  Equation 26
 
State 3: ( RF   pos   −RF   neg )×( LO   90   −LO   270 )× A 1 e   jP1   +DC 1  Equation 27
 
State 4: ( RF   pos   −RF   neg )×( LO   90   −LO   270 )× A 1 e   jP1   +DC 1  Equation 28
 
   A higher-order delta-sigma modulator  321  running off a clock  322  higher than the RF signal  300  modulation bandwidth is used to choose the LOPI and BBDI state. The delta-sigma modulator  321  generates a pseudo-random number from 1 to 4. As an example, a Bluetooth™ signal&#39;s bandwidth is 1 MHz, thus the interleaving may occur faster than 1 MHz such as 10 MHz. To continue the example, the delta-sigma pseudo random number modulator  321  would generate 10 million random numbers per second; these numbers generated from the set 1, 2, 3 and 4. Delta-sigma pseudo random numbers force the switching noise to higher frequencies. As an option, the numbers may be completely random. 
   For the in-phase signal component, the DC offset has 4 values DC 1 , −DC 1 , DC 2 , −DC 2  corresponding to states  1 ,  2 ,  3 ,  4  respectively. Likewise the quadrature baseband signal has  4  DC offset values each corresponding to a different interleaving combination. The amplitude and phase imbalance distortion has two values for the in-phase baseband signal: A 1 e jP1  for states  1  and  2  and A 2 e jP2  for states  3  and  4 . The quadrature baseband signal follows the opposite pattern for amplitude and phase imbalance. 
   Since a low pass filter can be interpreted as a time averaging function, the DC offset introduced by the mixers is averaged out in the baseband signals. With equal numbers of switching matrix states occurring, the DC offset is removed. This is summarized in the Equation 29.
 
 DC   M1   −DC   M1   +DC   M2   −DC   M2 =0  Equation 29
 
   Similarly, the in-phase and quadrature baseband signal imbalances average to the same value denoted in Equations 30 and 31.
 
 BB   I   =RF×LO   I ×( A 1 e   jP1   +A 1 e   jP1   +A 2 e   jP2   +A 2 e   jP2 )  Equation 30
 
 BB   Q   =RF×LO   Q ×( A 1 e   jP1   +A 1 e   jP1   +A 2 e   jP2   +A 2 e   jP2 )  Equation 31
 
   So while the amplitude and phase distortion are still present in the I and Q baseband signals, the distortion is now equal in the I and Q baseband signals. Since the signals are balanced, the amplitude and phase distortion does not degrade the system performance. With only the addition of a complex multiplicative term, Equations 30 and 31 are identical to Equations 1 and 2 which are the expressions for an ideal direct-conversion receiver. 
   The present technology thus provides a solution for important drawbacks of a direct-conversion receiver: DC offset and quadrature imbalance. 
     FIG. 6  is an illustration of a multi-mode wireless communication system in which one embodiment may be used. It should be understood that the components shown in  FIG. 6  are merely representative of one mode of wireless communication system and that other communication systems may use different components in order to achieve similar, or even different results. For example, a wired transceiver communication system may also be employed. The claims, therefore, are not intended to be limited to the system shown in  FIG. 6 . For example, the present technology may be implemented in a single-mode system. 
   In the wireless communication system of  FIG. 6 , multi-mode, wireless communication devices, otherwise referred to herein simply as wireless communication devices, are shown as wireless communication devices  100   a ,  100   b , and  100   n , one or more wireless communication devices being assigned to each user in the system. The designations a, b, and n on the wireless communication device identifiers correspond respectively to a first user, a second user, and an nth user, representing “n” number of users in the communication system. Although only three wireless communication devices  100  are shown in  FIG. 6 , it should be understood that a wireless communication system typically comprises many thousands of users. 
   Referring again to  FIG. 6 , control station  120  typically includes interface and processing circuitry for providing system control to base stations  110   a  through  110   n , representing one through “n” base stations comprising the wireless communication system. Base stations are provided for transmitting and receiving communication signals to and from wireless communication devices. Each base station  110  provides a coverage area ranging up to several miles in radius from the base station location. As wireless communication devices travel within the coverage area of each base station, communication signals to be transferred to and from the wireless communication device are routed generally through the particular base station to which the wireless communication device is most closely located. 
   Control station  120  provides circuitry for routing communications between wireless communication devices operating in various base station coverage areas, as well as between remote stations and land-line telephone users through a Public Switch Telephone Network, shown in  FIG. 6  as the PSTN  130 . Control station  120  may, alternatively, or in addition to, be connected to computer network  160  to provide communications between wireless communication devices in the communication system and various known computing devices connected to computer network  160 , such as personal computers, mainframe computers, digital cameras, email systems, remotely controlled devices, and so on. 
   Control station  120  typically comprises a telecommunications switch (not shown) and a Base Station Controller (BSC) (also not shown). The telecommunication switch provides a switching interface to PSTN  130  while the BSC provides the necessary hardware and software for communications to take place between base stations. Control station  120  provides other functions in the communication system as well, such as billing services and data services. 
   Control station  120  may be coupled to the base stations by various means such as dedicated telephone lines, optical fiber links, or microwave communication links. When a call is initiated by a wireless communication device, a paging message is transmitted to one or more base stations proximate to the wireless communication device initiating the call, generally over a paging channel. The paging message is routed to control station  120 , where it is processed and routed either to PSTN  130  or to one or more base stations proximate to a wireless communication device for which the call is intended. When a call is initiated from PSTN  130 , a paging message is received by control station  120  where it is then converted into a format suitable for the particular wireless communication system. 
   In the exemplary embodiment, the wireless communication device  100  is able to communicate in at least two modes, or types, of communications, data communications and voice communications. Data communication mode is used when it is desirous to send or receive information generally suitable for digital computational devices, such as laptop computers. Data is generally transmitted in discreet segments called packets. Each data packet generally contains overhead information used for a variety of purposes. For example, many data packets contain a data field used to store an error detection code. The error detection code may be used to check a received data packet to ensure that it was received intact; that is, the data was not corrupted during the transmission process. 
   Voice communication mode is used when it is desirous to transmit acoustic information, including human speech, facsimile tones, music, or other audible forms of communication. In voice communication mode, audio information is transmitted using one or more well-known wireless communication modulation techniques, such as CDMA, TDMA, AMPS, and others. 
   During typical voice communications, an over the air channel is established between one or more base stations and a wireless telephone. The channel is maintained throughout the duration of the voice call, no matter how much or little voice activity is occurring between the wireless telephone and the base station. In many instances, voice data is digitized and formatted into packets prior to transmission. Voice packets differ from data packets in that no information as to a destination address is contained within the voice packets. That is, a connection is first established between two locations, then voice data is transmitted between the two locations. No address information need be contained within the voice packets as the source and destination of the voice packets are predetermined by the connection. 
   Data mode may further include a capability of transmitting voice in certain applications. In this scenario, voice is digitized using techniques well known in the art. The digitized voice signals may be encrypted to provide for secure voice transmissions over the air. The digitized voice signals are then formatted into data packets, which are then transmitted over the air using well-known data transmission protocols. As explained above, each data packet contains information as to the address, or destination, of where the data packet is to arrive. 
     FIG. 7  illustrates the functional components of a wireless communication device, or wireless communication device,  100 , shown in block diagram format. It should be understood that the components shown in  FIG. 7  are merely representative of one mode of wireless communication device and that other communication devices may use different components in order to achieve similar, or even different results. The claims, therefore, are not intended to be limited to the system shown in  FIG. 7 . 
   Wireless communication device  100  is capable of multi-mode communications, meaning that it can operate in several modes of communications, such as voice communications or data communications. It should be understood that voice communications comprise any audio information including speech, music, or audible tones used for call processing, modems, and facsimile machines. Data communications comprise synchronous or asynchronous data transmission. In addition to these modes, wireless communication device is also capable of other modes of communications as well. 
   A user of wireless communication device  100  initiates communications generally by using input device  200 . Input device  200  comprises a keypad in the exemplary embodiment, however, input device  200  could be any device which accepts user commands, such as a voice response device which converts voice commands into electrical signals suitable for processing by controller  202 . During voice communications, the user speaks into microphone  204 , which transforms acoustic energy into electrical energy and sends the electrical signals to controller  202  for processing. 
   Microphone  204  may be substituted for input device  200  in an application where a second audio input device is undesirable. In many instances, a voice encoder/decoder, generally known as a Codec, is used between microphone  204  and controller  202 , or is incorporated within controller  202 , to convert the electrical signals from microphone  204  into a format more suitable for transmission over a limited bandwidth air interface. 
   Speaker  206  is used to convert received electrical signals into acoustic signals. Speaker  206  may comprise a speaker suitable for low volume acoustic outputs, typically for use in a traditional telephone application, or speaker  206  may comprise a loudspeaker, suitable for high volume acoustic outputs, typically for use in a dispatch applications. In another embodiment, speaker  206  may comprise a combination of the high volume and low volume acoustic speakers. 
   Wireless communication device  100  further comprises display  208  for allowing a user to view operational characteristics of the wireless communication device. Such displays are common in many of today&#39;s wireless devices including telephones and remote data terminals. 
   Data port  210  serves as an interface between controller  202  and external hardware devices. Data port  210  generally allows a variety of bi-directional data communications to take place between wireless communication device  100  and the external device. Such external devices include laptop computers, facsimile machines, and remote data terminals, among others. 
   When a user initiates voice or data communications, an identification code corresponding to a second communication device, generally a telephone number, is entered using input device  200 . In the exemplary embodiment, input device  200  comprises keys corresponding to digits 0 through 9, as well as additional function keys, such as SEND, END, and so forth. Input device  200  may also comprise one or more keys used to classify an outgoing communication as being a data communication or a voice communication. 
   For example, a user wishing to initiate a data communication might press a key designated for data communications, then dial a telephone number corresponding to a data device that the user wishes to communicate with. In one embodiment, all calls from wireless communication device  100  are assumed to be voice calls, unless classified as some other mode of communication, as described by one of the methods above. 
   Controller  202  serves as the main computational unit of wireless communication device  100 . Although controller  202  is shown as a single element in  FIG. 7 , it should be understood that controller  202  may comprise one or more individual components such as one or more Application Specific Integrated Circuits (ASICs) in combination with memory devices, bus controllers, and other support devices well known to those skilled in the art. 
   To facilitate the transmission and receipt of wireless RF signals in the foregoing context, an RF transceiver  212  and an antenna  214  are coupled to controller  202  for sending and receiving such signals. Similar to the controller  202 , one or more ASICs in combination with memory devices, bus controllers, etc. may be used to provide the RF transceiver  212 . Moreover, the aforementioned direct-conversion receiver may be incorporated into the RF transceiver  212  and/or controller  202  in any desired capacity for providing an improved system. 
   Working in conjunction with the controller  202  is memory  216 . The memory  216  is a device used to store information represented in digital format. Examples of memory  216  include random access memory (RAM), electrically erasable programmable read-only memory (EEPROM), non-volatile memory, and other known storage devices. 
   While various embodiments have been described above, it should be understood that they have been presented by way of example only, and not limitation. Thus, the breadth and scope of a preferred embodiment should not be limited by any of the above-described exemplary embodiments, but should be defined only in accordance with the following claims and their equivalents.