Abstract:
A spread spectrum noise shaper uses a modulation technique to achieve a greater signal-to-noise or signal-to-interference ratio (SNR or SIR). The technique doubles the system SIR, in principle. This doubling yields a doubling in system capacity. SNR is increased by receiving the spread spectrum signal in the presence of less noise near the edge of the spread spectrum bandwidth. The technique requires only small additions to a conventional spread spectrum system, in the form of an extra modulator at the transmitter, and an extra demodulator and filter at the receiver.

Description:
CROSS REFERENCE TO RELATED APPLICATION 
   This application claims priority from U.S. patent application Ser. No. 10/029,707, which was filed on Dec. 21, 2001, now U.S. Pat. No. 6,654,407, which is a continuation of U.S. patent application Ser. No. 09/679,773, which was filed on Oct. 5, 2000, now U.S. Pat. No. 6,347,111, which is a continuation of U.S. patent application Ser. No. 09/192,703, which was filed Nov. 16, 1998, now U.S. Pat. No. 6,148,021, which is a continuation of U.S. patent application Ser. No. 08/910,069, which was filed Aug. 12, 1997, now U.S. Pat. No. 5,838,719, which is continuation of U.S. patent application Ser. No. 08/542,306, which was filed on Oct. 12, 1995 (now abandoned), which is a continuation of U.S. patent application Ser. No. 08/146,650, which was filed on Nov. 2, 1993, now U.S. Pat. No. 5,459,758, which is incorporated by reference as if fully set forth herein. 

   FIELD OF THE INVENTION 
   The present invention relates to the field of spread spectrum or Code Division Multiple Access (CDMA) communications. 
   BACKGROUND OF THE INVENTION 
   In any communication system, it is desirable to increase the Signal to Noise Ratio (SNR) of the system. This has the advantageous effect of increasing the fidelity of a system for transmitting an analog signal, or decreasing the bit error rate of a system for transmitting a digital signal. Prior art methods for increasing the SNR of a system focus on increasing the available signal power at the input to the receiver, or decreasing the amount of noise in the receiver. The prior art teaches several ways to increase the signal power at the receiver, for instance, one may increase the gain of the RF amplifier in the transmitter, or make antennas larger or more directive. Another way to increase the effective power at the receiver, in the prior art, is to increase the bandwidth used to transmit the signal by spectrum spreading, or less directly, by coding for error correction. Spread spectrum techniques are taught in the books  Principles of Communication Systems , Second Edition, by Herbert Taub and Donald L. Schilling, McGraw Hill, 1986, and  Spread Spectrum Systems , Second Edition, by Robert C. Dixon, John Wiley &amp; Sons, 1984. One prior art technique for decreasing the amount of noise in the receiver is that of using a receiver with a low-noise amplifier at the front end of the receiver. 
   Spread spectrum communication systems typically operate with a very low power density spread over a wide enough bandwidth, sometimes known as the chipping bandwidth, to achieve a certain processing gain and, hence, required SNR for a given communication task. The noise in a spread spectrum system is often largely self-interference, as well as additive Gaussian white noise, which appear in the receiver as an interfering random signal having maximum power in the center of the spread spectrum bandwidth, in the portion of that bandwidth in which the desired spread spectrum signal is conventionally received. It is frequently desirable to increase the SNR in a system, especially if this can be achieved with only minimal changes to existing hardware, for instance, without changing antennas or RF amplifiers, and without significantly increasing the power or power density or occupied bandwidth of the transmitted signal. Increasing the SNR of a system significantly increases the capacity of the system. A 3 dB increase in SNR for a system allows an approximate doubling of the number of users that can be supported by the system. 
   SUMMARY OF THE INVENTION 
   The present invention is an improvement of a conventional spread spectrum communication system, having a transmitter and a receiver. It is an aspect of the present invention to increase the signal to noise ratio of a spread spectrum system by up-converting, or equivalently, frequency translating the signal, or up-shifting the spectrum of the signal to be sent, before spreading. Spreading, as used here, means spread spectrum encoding. In the present invention, the up-converted signal is transmitted and then received and then down-converted, after despreading. This has the advantageous effect of reducing the effect of self-interference and noise when the signal is received, when using a receiver according to the principles of the present invention. 
   In the present invention, the spectrum of the signal to be transmitted is shifted or frequency translated to the edge of the spread spectrum bandwidth, where noise in the receiver is less than in the middle of that bandwidth, prior to transmission. The signal is then received, in the signal processing sense, near the edge of the spread spectrum bandwidth, and then the spectrum of the received signal is frequency translated, or equivalently, retranslated, in the receiver. In one embodiment of the invention, these frequency translations are performed by adding a Single Sideband (SSB) modulator to the transmitter and an SSB demodulator to the receiver. This allows reception of the desired signal in a position near the edge of the chipping bandwidth that is less noisy than the position that is conventionally used for signal reception. The present invention allows reception of the same signal level as in a conventional system, in the presence of a lower level of received noise. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a block diagram of a conventional spread spectrum transmitter; 
       FIG. 1A  is a timing diagram of a pseudonoise (PN) sequence used in  FIGS. 1 ,  3 ,  5 , and  7 ; 
       FIG. 2A  is the power spectrum of the PCM data sequence of  FIG. 1 ; 
       FIG. 2C  is the power spectrum of the PN sequence of  FIG. 1 ; 
       FIG. 2D  is the power spectrum of the output of the spread spectrum encoder of  FIG. 1 ; 
       FIG. 3  is a block diagram of one embodiment of the spread spectrum transmitter of the present invention; 
       FIG. 3A  is a block diagram of the single-sideband modulator of  FIG. 3 ; 
       FIG. 4A  is the power spectrum of the PCM data sequence of  FIG. 3 ; 
       FIG. 4B  is the power spectrum of the output of the SSB modulator of  FIG. 3 ; 
       FIG. 4C  is the power spectrum of the PN sequence of  FIG. 3 ; 
       FIG. 4D  is the power spectrum of the output of the spread spectrum encoder of  FIG. 3 ; 
       FIG. 5  is a block diagram of a conventional spread spectrum receiver; 
       FIG. 6A  is the power spectrum of the received and filtered Intermediate Frequency (IF) signal of  FIG. 5 ; 
       FIG. 6B  is the power spectrum of the PN sequence of  FIG. 5 ; 
       FIG. 6C  is a power spectrum of the PCM data sequence component of the output of the spread spectrum decoder of  FIG. 5 ; 
       FIG. 6D  is a power spectrum of the noise component of the output of the spread spectrum decoder of  FIG. 5 ; 
       FIG. 6E  is the power spectrum of the combined signal and noise outputs of the spread spectrum decoder of  FIG. 5 ; 
       FIG. 6G  is the power spectrum of the PCM output of  FIG. 5 ; 
       FIG. 7  is a block diagram of one embodiment of the spread spectrum receiver of the present invention; 
       FIG. 8A  is the power spectrum of the received and filtered Intermediate Frequency (IF) signal of  FIG. 7 ; 
       FIG. 8B  is the power spectrum of the PN sequence of  FIG. 7 ; 
       FIG. 8C  is a power spectrum of the PCM data sequence component of the output of the spread spectrum decoder of  FIG. 7 ; 
       FIG. 8D  is a power spectrum of the noise component of the output of the spread spectrum decoder of  FIG. 7 ; 
       FIG. 8E  is the power spectrum of the combined signal and noise outputs of the spread spectrum decoder of  FIG. 7 ; 
       FIG. 8F  is the power spectrum of the output of the intermediate band-pass filter of  FIG. 7 ; and 
       FIG. 8G  is the power spectrum of the PCM output of  FIG. 7 ; 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1  shows a typical prior art spread spectrum transmitter. At the left-hand side of  FIG. 1 , a baseband, pulse coded modulated signal, typically a 32 kilobit per second Adaptive Pulse Code Modulated (ADPCM) speech signal, is applied to the left-hand terminal of mixer  104 , which is shown contained within spread spectrum encoder  105 . More information about the use and characteristics of spread spectrum encoders may be found in Taub, op. cit., pages 721–727. A pseudonoise (PN) sequence ( FIG. 1A ) is applied to the upper terminal of mixer  104 . Mixer  104  thereby performs a frequency spectrum spreading function by multiplying the PCM data sequence by the PN sequence in the time domain, which is equivalent to convolving the bimodal spectrum of the data sequence with the approximately rectangular spectrum of the PN sequence in the frequency domain. The output of mixer  104  is applied to low-pass filter  106 , whose cutoff frequency is equal to the system chip rate, Fcr. The output of filter  106  is then applied to one terminal of mixer  108  and suitably up-converted, as determined by the carrier frequency Fc, applied to its other terminal. The up-converted signal is then passed through bandpass filter  110  (typically a helical resonator), which has a bandwidth equal to twice the chip rate and a center frequency equal to the center frequency of the spread spectrum system&#39;s channel bandwidth. The output of filter  110  is applied to the input of broadband RF amplifier  112 , whose output drives antenna  114 . 
     FIG. 1A  shows a conventional spreading sequence, which is a pseudo-random digital sequence, in a typical embodiment of this invention. It is conventionally shown as attaining two constant values over time, the values of ±1. The sequence of  FIG. 1A  is used to spread the signal to be transmitted and to despread the received signal. Information as to the generation and use of PN sequences may be found in Taub, op. cit., pages 732–737. 
     FIGS. 2A ,  2 C and  2 D illustrate power spectra in the prior art transmitter of  FIG. 1 .  FIG. 2A  shows the power spectrum of the PCM data sequence of  FIG. 1 . It is shown as essentially bandlimited to Fbr, the bit rate of the data sequence.  FIG. 2C  shows the power spectrum of the PN sequence used in  FIG. 1 .  FIG. 2D  shows the power spectrum of the output of mixer  104  of  FIG. 1 , which is up-converted and then band-pass filtered by band-pass filter  110 , as indicated by the vertical lines labeled −Fcr and +Fcr of  FIG. 2D . 
     FIG. 3  illustrates a block diagram of one embodiment of the spread spectrum transmitter of the present invention. Referring to  FIG. 3 , in accordance with the principles of the present invention, the baseband PCM-signal of  FIG. 3  is first passed through a pre-modulation stage, comprising single-sideband modulator  302 . If modulator  302  were omitted,  FIG. 3  would show a conventional spread spectrum transmitter of the prior art, i. e.  FIG. 1 . Modulator  302  is supplied with a spectrum shifting signal whose frequency determines how far apart the two halves of the bimodal data spectrum ( FIG. 4A ) of the data sequence are to be frequency translated or shifted. The shifted bimodal data spectra are illustrated in  FIG. 4B . In one embodiment of the invention, the reference signal input to modulator  302  has a constant frequency of Fcr, typically 10 MHz. Fcr is called the chip rate of the spread spectrum communication system. The details of modulator  302  are further described below, in conjunction with  FIG. 3A . The output of modulator  302  is applied to the left hand terminal of mixer  304 , which is shown contained within spread spectrum encoder  305 . The upper terminal of mixer  304  is supplied with a PN sequence as shown in  FIG. 3 . The output of mixer  304  is low-pass filtered in filter  306 , which has a cutoff frequency equal to Fcr, the system chip rate. The output of filter  306  is subsequently up-converted in mixer  308 , using a suitable carrier reference frequency, which is approximately the same as the carrier reference frequency in mixer  706  of  FIG. 7 . 
     FIG. 3A  illustrates single-sideband modulator  302 . The PCM data sequence from  FIG. 3  is converted into I and Q components by phase shift network  318 . The output of network  318  drives balanced modulators  320  and  322 , which are respectively fed with sinusoidal and cosinusoidal carrier signals from phase shift network  324 . The outputs of modulators  320  and  322  are combined by adder  326 , whose output is the output of modulator  302 . 
     FIGS. 4A ,  4 B,  4 C and  4 D illustrate power spectra in the transmitter of a typical embodiment of the present invention, i. e. the transmitter of  FIG. 3 .  FIG. 4A  shows the power spectrum of the PCM data sequence of  FIG. 3 . It is shown as essentially bandlimited to Fbr, the bit rate of the PCM data sequence.  FIG. 4B  shows the power spectrum of the output of SSB modulator  302  of  FIG. 3 ,  FIG. 4C  shows the power spectrum of the PN sequence used in  FIGS. 3 and 7 .  FIG. 4D  shows the power spectrum of the output of mixer  304  of  FIG. 3 . 
     FIG. 5  shows a conventional, or prior art spread spectrum receiver. Antenna  502  of  FIG. 5  receives a spread spectrum signal, which is filtered by filter  504 , which has a bandwidth equal to twice the chip rate, and a center frequency equal to the center frequency of the spread spectrum system&#39;s channel bandwidth. The output of filter  504  is subsequently down-converted by mixer  506 , possibly in two stages, to a baseband signal, using a local oscillator having a constant frequency which is approximately the same as the carrier frequency Fc of  FIG. 1 . The output of mixer  506  is then despread, or equivalently, spread spectrum decoded, by applying it to the left hand terminal of mixer  508 , while applying the PN sequence of  FIG. 1A  to the upper terminal of mixer  508 . Mixer  508  is shown as contained within spread spectrum decoder  509  of  FIG. 5 . The use and characteristics of spread spectrum decoders is discussed more fully in the book by Taub. More particularly, it will be appreciated that the PN sequence used in the receiver of a spread spectrum communication system must be synchronized with the PN sequence used in the transmitter. Methods for achieving this synchronization are discussed in pages 744–748 of Taub, op. cit. The output of mixer  508  is applied to low-pass filter- 514 , which has a cutoff frequency at the data rate Fb of the PCM data sequence of  FIG. 1 . The output of filter  514  is a replica of the PCM data sequence of  FIG. 1 . 
     FIGS. 6A ,  6 B,  6 C,  6 D,  6 E and  6 G show the operation of a conventional spread spectrum receiver.  FIG. 6A  illustrates the power spectrum of the received signal+noise, as it would be measured at the output of band-pass filter  504  of  FIG. 5 .  FIG. 6B  illustrates the power spectrum of the despreading sequence of  FIG. 5 .  FIG. 6C  illustrates the power spectrum of the PCM data sequence of  FIG. 1 , after it is spread spectrum decoded, i. e. despread, using the sequence of  FIG. 1A . 
     FIG. 6D  illustrates the power spectrum of the noise component of the output of the correlator of a spread spectrum receiver, as might be measured at the output of mixer  508  of  FIG. 5 . The triangular shape of the noise component shown in  FIG. 6D  corresponds to the convolution in the frequency domain of the two approximately rectangular spectra of  FIGS. 6A and 6B . It also corresponds to the multiplication of these two signals in the time domain.  FIG. 6D  shows that noise or self-interference that is uncorrelated with the spreading sequence used in the receiver appears in the receiver as a triangular power spectrum with a maximum magnitude at the center of the spread-spectrum bandwidth, as translated to baseband, and an amplitude of zero at twice the spread spectrum bandwidth. In a conventional spread spectrum receiver, the desired signal (as shown in  FIG. 6C ) is received at the center of the spread spectrum bandwidth, where the noise component ( FIG. 6D ) has a maximum magnitude. 
     FIG. 6E  illustrates the power spectrum of the combined data sequence+noise output of mixer  508  of  FIG. 5 , after low-pass filtering.  FIG. 6E  represents the superposition of  FIGS. 6C and 6D .  FIG. 6G  shows the result of low-pass filtering the spectrum of  FIG. 6E , i. e.  FIG. 6G  shows the output of a conventional spread spectrum receiver, for comparison with the output of one embodiment of the receiver of the present invention. 
     FIG. 7  shows one embodiment of the spread spectrum receiver of the present invention.  FIG. 7  differs from the conventional spread spectrum receiver of  FIG. 5  in that filter  710  and SSB demodulator  712  have been added to  FIG. 5  to produce  FIG. 7 . Antenna  702  of  FIG. 7  receives a spread spectrum signal, which is filtered by filter  704 , which has a bandwidth equal to twice the chip rate and a center frequency equal to the center frequency of the spread spectrum system&#39;s channel bandwidth. The output of filter  704  is then down-converted by mixer  706 , possibly in two stages, to a baseband signal, using a local oscillator having a constant frequency which is approximately the same as the carrier frequency of  FIG. 3 . The output of mixer  706  is then despread with mixer  708 , using the PN sequence of  FIG. 1A . The output of mixer  708  is then filtered in band-pass filter  710 , which has a bandwidth equal to the data rate and a center frequency equal to the system chip rate. The output of filter  710  undergoes single-sideband demodulation in SSB demodulator  712 , using a reference signal having a constant frequency equal to the chip rate. The output of demodulator  712  is low-pass filtered in filter  714 , which has a cutoff frequency at the data rate of the PCM data sequence of  FIG. 3 . The output of filter  714  is a replica of the PCM data sequence of  FIG. 3 . 
   The present invention resides in the processing steps illustrated in  FIGS. 8E ,  8 F, and  8 G.  FIG. 8A  illustrates the power spectrum of the received signal+noise, as it would be measured at the output of band-pass-filter  704  of  FIG. 7 .  FIG. 8B  illustrates the power spectrum of the despreading sequence of  FIG. 7 .  FIG. 8C  illustrates the power spectrum of the PCM data sequence of  FIG. 3 , after it is spread spectrum decoded, i. e. despread, using the sequence of  FIG. 1A . 
     FIG. 8D  illustrates the power spectrum of the noise component of the output of the spread spectrum decoder or correlator of a spread spectrum receiver, as might be measured at the output of mixer  708  of  FIG. 7 . The triangular shape of the noise component shown in  FIG. 8D  corresponds to the convolution in the frequency domain of the two approximately rectangular spectra of  FIGS. 8A and 8B . It also corresponds to the multiplication of these two signals in the time domain.  FIG. 8D  shows that noise or self-interference that is uncorrelated with the spreading sequence used in the receiver appears in the receiver as a triangular power spectrum with a maximum magnitude at the center of the spread-spectrum bandwidth, as translated to baseband, and an amplitude of zero at twice the spread spectrum bandwidth. In a conventional spread spectrum receiver, the desired signal (as shown in  FIG. 6C ) is received at the center of the spread spectrum bandwidth, where the noise component ( FIG. 8D ) has a maximum magnitude. 
     FIG. 8E  illustrates a portion of the power spectrum of the combined data sequence+noise output of mixer  708  of  FIG. 7 .  FIG. 8E  represents the superposition of  FIGS. 8C and 8D .  FIG. 8E  illustrates the result of mixing the sequence of  FIG. 1A  with the output of down converter  706 , in accordance with the present invention. It will be noted that, compared to  FIG. 6E ,  FIG. 8E  shows that the two halves of the spectrum of the PCM data sequence are separated by twice the spread spectrum bandwidth. It will be further noted that the noise of  FIG. 6E , representing the output of a conventional spread spectrum receiver, is at a maximum near the single copy of the data spectrum, at the center frequency, while the noise in  FIG. 8E  is reduced in intensity to approximately half that of the noise of  FIG. 6E , in the vicinity of the data spectra. 
     FIG. 8F  illustrates the result of band-pass filtering the spectrum illustrated in  FIG. 8E , using band-pass filter  710 .  FIG. 8G  shows the result of single-sideband demodulating the two halves of the PCM data spectrum of  FIG. 8F  in demodulator  712 , using a reference signal having a constant frequency equal to that of the system chip rate.  FIG. 8G  may be used to compare the signal to noise ratio of the present invention to that of a conventional spread spectrum receiver.  FIG. 6E  shows an SNR of 2:1, while  FIG. 8E  shows an SNR of 4:1, near each half of the spectrum of the PCM data sequence. This is also the case in  FIG. 8F .  FIG. 8G  shows that translating the two halves of the PCM data spectrum of  FIG. 8F  by single-sideband demodulation produces a signal amplitude of 4, as in  FIG. 6E . Demodulating the two noise components of the spectra of  FIG. 8F  shifts those noise components to the center of the spectrum without increasing their magnitude. The amplitude of the noise component of the spectrum of the output of demodulator  712 , after low-pass filtering in filter  714 , is approximately 1, as illustrated in  FIG. 8G . Thus  FIG. 8G  shows a 4:1 signal to noise ratio, using the present invention, which is a 3 dB improvement in the signal to noise ratio of the received signal, compared to the SNR of 2:1 illustrated in  FIG. 6G , which corresponds to the output of a conventional spread spectrum receiver. 
   While the above description contains many specifics, these should not be construed as limitations on the scope of the invention, but rather as examples of several of the many possible embodiments. Many other variations are possible. For example, this method of spread spectrum modulation could be used in any medium in which spread spectrum is employed, not just radio. Digital modulation techniques may also allow the performance gain described above. One such digital technique uses a modified spreading sequence in the transmitter, in which the despreading sequence of the receiver is multiplied by a digital sinusoid (the sequence 1010101010) at the system chip rate. The receiver then despreads with the original despreading sequence, band-pass filters and coherently combines the resulting spectra, multiplying the band-passed signal by the digital sinusoid.