Abstract:
In digital signal recording apparatus using I-NRZI modulation for recording, the need for intermittently reading or intermittently writing buffer storage is eliminated by using parallel-bit precoding to generate the channel words that are selected between for recording. The precoders perform preceding on an accelerated basis using ripple-through integration of the alternate successive bits used to form each channel word. Two precoders generate (n+1)-parallel-bit channel words at a channel word rate slower by a factor of (n+1) than the rate of a system clock for the I-NRZI modulation. This leaves additional time during each channel word interval to carry out a decision procedure, which determines which of the channel words generated by the two precoders is to be selected for recording. There is also sufficient additional time for completing a subsequent updating procedure, in which precoding information stored in the precoder that did not generate the selected channel word is altered, to conform to preceding information stored in the precoder that did generate the selected channel word. The parallel-bit channel words from the precoders are converted to serial-bit format for recording with a bit rate equal to that of the system clock. The parallel-bit channel words from the precoders are converted to serial-bit format with an effective bit rate that is substantially higher than that of the system clock, to provide signal for timely implementing the decision and updating procedures.

Description:
The present invention relates to digital signal recording apparatus recording interleaved non-return-to-zero, invert-on-ONEs (I-NRZI) modulation that includes pilot signals used for head tracking during playback. 
     BACKGROUND OF THE INVENTION 
     In a magnetic recording/reproducing apparatus such as a videocassette recorder, as a head deviates from a track on a magnetic recording medium during playback, head output is decreased and errors increase. This precludes the normal reproduction of an image, so it is required for the head to trace a target track precisely. In other words, it is necessary to maintain head tracking. In order to extend recording time in a digital videocassette recorder for home use, tracks are especially narrow, which increases the precision of the head tracking needed for satisfactory reproduction of images. Among the methods for detecting head tracking error, or deviation from ideal tracking, are methods that use different respective pilot signals for successive tracks to facilitate comparison of the crosstalk of the pilot signals from the tracks preceding and succeeding the track being most closely followed by the head, thus to detect whether the head tracking deviates toward the preceding track or toward the succeeding track. The pilot signals take the form of peaks and notches in the frequency spectra of the digital signals recorded on the tracks by selecting between two types of interleaved non-return to-zero, invert-on-ONEs (I-NRZI) modulation. The same information is encoded into two parallel-in-time sets of serially supplied channel words; and the channel words that are selected from one or the other of the sets to control I-NRZI modulation during recording, are selected so the I-NRZI modulation will deviate least from the pilot signal criterion for each recording track. When the selection of the channel word is completed, preceding information stored in the precoder that did not generate the selected channel word is altered, to conform to preceding information stored in the precoder that did generate the selected channel word. This is done to provide continuity of the preceding procedures and of the decoding procedures subsequent to the I-NRZI modulation being recovered from the recording medium during playback and demodulated. When the selection of the channel word is completed, integrators in the circuitry for determining which channel word is to be selected have to have their contents updated to reflect which channel word was in fact selected for recording. Such methods are described in U.S. Pat. No. 5,142,421 issued Aug. 25, 1992 to Kahlman et alii, entitled “ DEVICE FOR RECORDING A DIGITAL INFORMATION SIGNAL ON A RECORD CARRIER ” and incorporated herein by reference. 
     In Kahlman et alii the generation of the I-NRZI modulation is done on a serial-bit basis. This does not lend itself to pipeline operation in which channel words selected from the serial-bit precoders are recorded on the magnetic recording medium, after some fixed delay to accommodate the selection circuitry. It takes some time after a pair of respective channel words are generated, for a decision procedure that determines which of them will be recorded. After the decision procedure, it then takes some further time for updating stored information in the precoders. These decision and updating procedures must be completed before further preceding is possible, so the delays caused by these decision and updating procedures introduce gaps into the continuous flow of bits as regularly clocked by synchronous clocking methods. The decision procedures have considerable delay time associated with them to permit digital multiplication, addition, integration and squaring procedures to be carried out, although squaring time can be reduced by using look-up tables stored in read-only memory. Accordingly, first-in/first-out buffer storage that can be intermittently read from has to be provided before the serial-bit precoders; and first-in/first-out buffer storage that can be intermittently written with the selected channel words and subsequently continuously read from has to be provided for channel words generated by the serial-bit precoders. The generation of clocking signals for the buffer storage is somewhat complex, so it is desired to avoid the need for intermittently written or intermittently read buffer storage. 
     SUMMARY OF THE INVENTION 
     In digital signal recording apparatus using I-NRZI modulation for recording, the need for intermittently reading or intermittently writing buffer storage associated therewith is eliminated by the invention. The preceding, used to generate the codes that control the generation of I-NRZI modulation, is performed on a serial-word, parallel-bits-per-word basis. The precoders are modified to perform precoding on an accelerated basis using ripple-through integration of the alternate successive bits used to form each channel word. Two precoders then generate (n+1)-parallel-bit channel words at a channel word rate slower by a factor of (n+1) than the rate of a system clock. This leaves additional time during each channel word interval to carry out a decision procedure that determines which of the channel words is to be selected for recording. There is also sufficient additional time to complete a subsequent updating procedure, in which preceding information stored in the precoder that did not generate the selected channel word is altered, to conform to precoding information stored in the precoder that did generate the selected channel word. The serial-word, parallel-bit channel words from the precoders that are selected for being recorded on the magnetic recording medium are converted to serial-bit format, with a bit rate equal to that of the system clock for the I-NRZI modulation being recorded. The serial-word, parallel-bits-per-word, codestreams from the precoders are converted to serial-bit format with an effective bit rate that is substantially higher than that of the system clock, to provide signal for implementing the decision and updating procedures in timely fashion. 
     In certain preferred embodiments of the invention the channel words from the precoders are each separated into two component subwords or divided-channel words when converted to serial-bit format, to form two parallel bitstreams each having a bit rate that is the same as the system clock used to control the I-NRZI modulation being recorded. The two parallel bitstreams provide input for the computations performed to determine which of the channel words is to be recorded, which input has an effective bit rate twice that of the system clock. 
     In alternative embodiments of the invention, in order to generate signals on which to base the computations performed for determining which of the channel words is to be recorded, channel words from the precoders are converted to serial-bit format having a bit rate that is actually twice the system clock rate used to control the I-NRZI modulation being recorded. The serial-bit channel words that have a bit rate twice system clock rate provide input for the computations performed to determine which of the channel words is to be recorded. 
    
    
     BRIEF DESCRIPTION OF THE DRAWING 
     FIG. 1 depicts an illustrative pattern for recording a serial data stream of channel words on adjacent parallel tracks within the surface of a magnetic recording medium. 
     FIGS. 2A,  2 B and  2 C illustrate frequency spectra associated with the patterns shown in FIG.  1 . 
     FIG. 3 is a block diagram of prior-art digital signal recording apparatus described in U.S. Pat. No. 5,142,421. 
     FIG. 4 is a detailed circuit diagram of a portion of an improved control signal generator for the digital signal recording apparatus shown in FIG. 3 which control signal generator generates selection signals for selecting channel words responsive to frequency spectrum analyses of proposed I-NRZ modulation performed digitally. 
     FIG. 5 illustrates one of the frequency spectra of the pattern of a serial data stream of channel words selected by a control signal generated from the control signal generator shown in FIG.  4 . 
     FIG. 6 is a block diagram of digital signal recording apparatus that embodies the invention. 
     FIG. 7 is a block diagram of other digital signal recording apparatus that embodies the invention. 
     FIG. 8 is a detailed block diagram of a portion of the FIG. 6 digital signal recording apparatus. 
     FIG. 9 is a detailed circuit diagram of the “0” bit inserter shown in FIG.  8 . 
     FIG. 10 is a detailed circuit diagram of the  2 T precoder shown in FIG.  8 . 
     FIG. 11 is a detailed circuit diagram of a parallel-to-serial converter shown in FIG.  8 . 
     FIG. 12 is a block diagram of the control signal generator used in the FIG. 6 digital signal recording apparatus. 
     FIG. 13 is a detailed circuit diagram of PATH 0  in a portion of the control signal generator shown in FIG.  12 . 
     FIG. 14A illustrates the waveform of a signal generated from the triangular wave generator shown in FIG.  13 . 
     FIG. 14B is a table of data stored in read-only memory (ROM) for implementing the triangular wave generator. 
     FIGS. 15A,  15 B and  15 C illustrate the sine-wave and square-wave signals used in FIG.  13 . 
     FIGS. 16A-16G illustrate operation waveforms for blocks of the diagram shown in FIG.  6 . 
     FIG. 17 is a detailed circuit diagram of PATH 0  in a portion of the control signal generator shown in FIG. 6 constructed in a way alternative to that shown in FIG.  13 . 
     FIG. 18 is a block diagram of still other digital signal recording apparatus that embodies the invention. 
     FIGS. 19A-19D illustrate operation waveforms of blocks of the diagram shown in FIG.  18 . 
     In the block diagrams, blocks with the legend “P/P” are parallel-to-parallel converters for converting consecutive serial groups of parallel-bit data each to parallel-bit words; blocks with the legend “P/S” are parallel-to-serial converters for converting parallel-bit data to serial-bit data; blocks with the legend “INT” are digital integrators; blocks with the legend “SQ” are digital squaring circuits; and blocks with the legend “L” are bit latches. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1 illustrates how, in order to facilitate head tracking, helical-scan digital recording apparatus records a serial data stream of channel words in three spectral response patterns F 0 , F 1  and F 2  on successive parallel tracks of a magnetic recording medium. Per convention, the tracks are shown shorter in length and more skewed from the direction of tape travel than is the actual case. Pilot signals appear in the spectra of digital signals recorded in the sequence of F 0 , F 1 , F 0 , F 2 , . . . on the respective tracks of the magnetic recording medium. The pilot signals take the form of notches or peaks at prescribed frequencies that are introduced into the frequency-domain spectral energy response (Fourier transform) of the signals recorded on the tracks. When playing back from any one of these tracks of a particular pattern, certain deviations of the frequency-domain spectral energy response from expected values is ascertained. Such deviations are ascribed to pick-up of the digital signals from the preceding and succeeding tracks, in order to estimate the relative proximity of the head to the preceding track and to the succeeding track, from which the tracking error of the head can be determined. The illustrated sequential pattern F 0 , F 1 , F 0 , F 2  is merely exemplary, since in practice the number of patterns and the recording sequence can be different from that of the illustration. U.S. Pat. No. 5,142,421 describes certain of these variants. 
     FIGS. 2A,  2 B and  2 C illustrate the frequency spectra of the serial-bit data streams of channel words bearing patterns F 0 , F 1  and F 2  shown in FIG. 1, respectively. In the frequency spectrum of the pattern F 0 , there are notches at frequencies f 1  and f 2  where the spectral energy is relatively small. In the frequency spectrum of the pattern F 1 , there is a pilot signal (peak) at frequency f 1 =ω 1 /2π where the spectral energy is relatively large, and there is a notch at frequency f 2 =ω 2 /2π where the spectral energy is relatively small. In the frequency spectrum of the pattern F 2 , there is a notch at frequency f 1  where the spectral energy is relatively small and a pilot signal (peak) at frequency f 2  where the spectral energy is relatively large. 
     During the playback of the pattern F 0 , a crosstalk effect between pilot signals (peaks f 1  and f 2 ) of the patterns F 1  and F 2  of adjacent tracks is used to determine tracking error. On one hand, if the head deviates from the center of the pattern F 0  toward the pattern F 1 , the crosstalk of pilot signal from the pattern F 1  becomes greater than that from the pattern F 2 . As a result, frequency component f 1  of a playback signal becomes greater and frequency component f 2  becomes smaller. On the other hand, if the head deviates from the center of the pattern F 0  toward the pattern F 2 , the crosstalk of pilot signal from the pattern F 2  becomes greater than that from the pattern F 1 . As a result, on average, frequency component f 2  of a playback signal becomes greater and frequency component f 1  becomes smaller. When playing back the pattern F 0 , then, comparing the average spectral energy of the playback signal at frequencies f 1  and f 2  enables the detection of deviation of head tracking. Using this result, precise tracking is made possible by controlling the height of a head element with a voltage applied to a piezo-electric element the head is mounted on, or by controlling the traveling speed of the magnetic recording medium (tape). 
     FIG. 3 is a block diagram of a digital signal recording apparatus disclosed in U.S. Pat. No. 5,142,421 issued Aug. 25, 1992 to Kahlman et alii, entitled “ DEVICE FOR RECORDING A DIGITAL INFORMATION SIGNAL ON A RECORD CARRIER ” and incorporated herein by reference. The schematic configuration and operation thereof will be described in regard to the conventional method for recording the patterns F 0 , F 1  and F 2 . 
     In FIG. 3, 8-parallel -bit digital words are supplied via an input port  1  to a parallel-to-serial (P/S) converter  2 . The P/S converter  2  converts, for instance, each succeeding group of three 8-parallel-bit digital words into a single 24-serial-bit digital information word supplied via a converter output port  3 . A signal inserting portion  4  includes a “0” bit inserter  4 . 1  and a “1” bit inserter  4 . 2  each receiving as respective input signal the stream of 24-serial-bit digital information words appearing at the output port  3  of the P/S converter  2 . The “0” bit inserter  4 . 1  inserts a single-bit digital prefix consisting of a “0” before the most significant bit (MSB) of each 24-serial-bit information word to generate a respective 25-serial-bit “positive” information word supplied from an output port  5  of the “0” bit inserter  4 . 1 . The “1” bit inserter  4 . 2  inserts a single-bit digital prefix consisting of a “1” before the most significant bit (MSB) of each 24-serial-bit information word to generate a respective 25-serial-bit “negative” information word supplied from an output port  7  of the “1” bit inserter  4 . 2 . 
     An encoder  6  includes a precoder  6 . 1  converting the “positive” information words to respective 25-serial-bit channel words supplied via a connection  9 . The encoder  6  further includes a precoder  6 . 2  converting the “negative” information words to respective 25-serial-bit channel words supplied via a connection  11 . In the remainder of this specification and in the claims appended to this specification, in order to distinguish between the channel words supplied from the precoder  6 . 1  and the channel words supplied from the precoder  6 . 2 , the channel words supplied from the precoder  6 . 1  are referred to as “positive”-information channel words; and the channel words supplied from the precoder  6 . 2  are referred to as “negative”-information channel words. If the precoders  6 . 1  and  6 . 2  are  2 T precoders, the single-bit prefix code causes them to generate two 25 serial-bit channel words in which the corresponding even bits are the same, and the corresponding odd bits are bit-complementary. A  2 T precoder comprises a two-input exclusive-OR gate and a two-stage shift register providing an integrating feedback connection from the output connection of the exclusive-OR gate to a first of its input connections. The exclusive-OR gate receives the precoder input signal at its second input connection, supplies the precoder output signal at its output connection, and normally receives at its first input connection the precoder output signal as delayed  2 T by passage through the two-stage shift register. The interval T is the sampling interval of the precoder input signal and the interval between clocked shifts of bits through the two-stage shift register. The feedback connection of the exclusive-OR gate provided by the two-stage shift register is referred to as the “integrating feedback connection” or simply the “integrating connection”. 
     The precoder  6 . 1  supplies 25-serial-bit “positive”-information channel words via the connection  9  as its output signal; and the precoder  6 . 2  supplies 25-serial-bit “negative”-information channel words via the connection  11  as its output signal. Based on these 25-serial-bit channel words supplied in parallel from the precoders  6 . 1  and  6 . 2 , a control signal generator  10  compares the respective frequency-domain spectral energy characteristics of each word to the prescribed spectral energy characteristics for the track that is to be recorded by a digital recorder  14  to determine which channel word deviates the least from the prescribed spectral response. The control signal generator  10  generates a control signal CS indicative of which of the channel words supplied from the precoders  6 . 1  and  6 . 2  deviates the least from the prescribed spectral response and should be selected for recording. Control signal CS is supplied via a connection  17  to the selection control port of a selector  12 , which selects the output signal from one of the precoders  6 . 1  and  6 . 2  (as delayed by a time compensator  8 ) that deviates the least from the prescribed spectral response, for application to the digital tape recorder  14 . Delays  8 . 1  and  8 . 2  of the time compensator  8  are needed to compensate for the time necessary for the control signal generator  10  to generate control signal CS for application to the selector  12 . The control signal CS is also supplied via the connection  17  to respective control ports of the precoders  6 . 1  and  6 . 2  to control the transfer of the contents of the shift register in the one of the precoders  6 . 1  and  6 . 2  the output from which is selected for recording to the shift register in the other of the precoders  6 . 1  and  6 . 2 , to provide for continuity of coding. 
     The selector  12  receives via a connection  13  the “positive”-information output of the precoder  6 . 1  as delayed by the delay  8 . 1  and receives via a connection  15  the “negative”-information output of the precoder  6 . 2  as delayed by the delay  8 . 2 . In response to the control signal CS the selector  12  supplies a selected one of the delayed output signals of the precoders  6 . 1  and  6 . 2  via a connection  19  to the digital recorder  14  as input signal for recording. Some rate buffering is required in order that the bit modulation can be recorded at a constant bit rate by the digital recorder  14 . The delays  8 . 1  and  8 . 2  can be fixed delays, with the rate buffering being provided after the selector  12 ; or, alternatively, the delays  8 . 1  and  8 . 2  may be first-in/first-out (FIFO) rate buffer memories that provide the necessary rate buffering in addition to always providing sufficient delay to complete the computations for deciding which of the output signals of the precoders  6 . 1  and  6 . 2  is to be recorded. 
     FIG. 4 is a detailed circuit diagram of an improved control signal generator for the FIG. 3 digital signal recording apparatus, as operated to generate a serial data stream of channel words with a frequency response spectrum as shown in FIG.  5 . As compared with the spectrum of pattern F 1  shown in FIG. 2B, in the spectrum shown in FIG. 5, dips occur on each side of f 1 . These dips indicate that the noise power of the spectrum is reduced next to the pilot signal frequency f 1 , which results in increased signal-to-noise ratio for the detection of pilot signal at frequency f 1 . 
     The improved control signal generator of FIG. 4 differs from that described by Kahlman et alii in that it performs frequency spectrum analyses in the digital regime, rather than in the analog regime per Kahlman et alii, and in that it accordingly includes code-to-arithmetic mappers  10 . 1  and  10 . 2 . The code-to-arithmetic mapper  10 . 1  converts the ONEs and ZEROs of the “positive”-information output of the precoder  6 . 1  to arithmetic descriptions of the I-NRZI modulation that switches between negative and positive arithmetic values of similar amplitude and is unaccompanied by a direct term. The code-to-arithmetic mapper  10 . 2  is similar in its construction to the code-to-arithmetic mapper  10 . 1 . The code-to-arithmetic mapper  10 . 2  converts the ONEs and ZEROs of the “negative”-information output of the precoder  6 . 2  to arithmetic descriptions of the I-NRZI modulation that switches between negative and positive arithmetic values of similar amplitude and is unaccompanied by a direct term. By way of example, each of the mappers  10 . 1  and  10 . 2  can use the ONEs and ZEROs supplied thereto as a changing sign bit before an unchanging ONE, so the modulation is described in two&#39;s complement arithmetic terms. 
     A sine/cosine look-up table stored in read-only memory (ROM), not shown, generates a complex carrier of frequency f 1 , having an angular frequency ω 1  and composed of sinω 1 t and cosω 1 t components. Another sine/cosine look-up table stored in ROM, not shown, generates a complex carrier of frequency f 2 , having an angular frequency ω 2  and composed of sinω 2 t and cosω 2 t components. A triangular wave generator  18  generates a triangular signal corresponding to a digital sum value of an intended frequency (f 1 ) of the serial data stream of channel words, and a square wave generator  38  generates a square wave of frequency f 1 . The triangular wave generator  18  and the square wave generator  38  can also be provided by look-up tables stored in ROM. The generation of all system functions in ROM simplifies processing the channel words in other than normal bit order. 
     Filter circuitry PATH 0  determines how the spectral energy distribution of the I-NRZI modulation, when the generation thereof continues based on a “positive”-information channel word from the precoder  6 . 1 , deviates from the desired spectral energy distribution for a track recorded with the F 1  pattern having a peak at frequency f 1 , a dip on either side of frequency f 1  and a notch at frequency f 2 . A weighted summation circuit  52 . 1  combines with appropriate weighting the computed deviation from the desired notch at zero frequency and the desired peak at frequency f 1 , as furnished from a squaring circuit  22 . 1 , with the computed deviations from the other desired features. The computed deviations from the notch at the frequency f 2 , as furnished for orthogonal phases of the frequency f 2  by squaring circuits  28 . 1  and  34 . 1 , are weighted similarly to each other in the weighted summation circuit  52 . 1 . The computed deviations from the dip on either side of frequency f 1 , as furnished for orthogonal phases of the frequency f 1  by squaring circuits  44 . 1  and  50 . 1 , are weighted similarly to each other in the weighted summation circuit  52 . 1 . The effective weighting of the inputs to the weighted summation circuit  52 . 1  from the squaring circuits  28 . 1  and  34 . 1  is relatively large compared to the weighting of the input to the weighted summation circuit  52 . 1  from the squaring circuit  22 . 1 , since lack of correct pilot frequency f 1  is better tolerated than presence of incorrect pilot frequency f 2  by the tracking correction circuitry used during playback. The effective weighting of the inputs to the weighted summation circuit  52 . 1  from the squaring circuits  44 . 1  and  50 . 1  is relatively small compared to the weighting of the input to the weighted summation circuit  52 . 1  from the squaring circuit  22 . 1 . The filter circuitry PATH 0  supplies, as the weighted sum output signal from the weighted summation circuit  52 . 1  therein, a first error signal e 1 . 
     The computation in the PATH 0  system of the amount by which the spectral energy distribution of the I-NRZI modulation, when the generation thereof continues based on a “positive”-information channel word from the precoder  6 . 1 , deviates from the desired notch at zero frequency and the desired peak at frequency f 1  peak is done in the following way. An integration circuit  16 . 1  receives the current “positive”-information channel word from the precoder  6 . 1 , as converted to arithmetic form by the code-to-arithmetic mapper  10 . 1 , and integrates it with a prestored value. A subtractor  20 . 1  subtracts the output signal of the triangular wave generator  18  from the output of the integration circuit  16 . 1 ; and a squaring circuit  22 . 1  for multiplies the resulting difference by itself; and the resulting square is supplied to the weighted summation network  52 . 1  to provide a component of the first error signal e 1 . The triangular wave generator  18  and the subtractor  20 . 1  provide detection circuitry for detecting any deviation from the prescribed digital sum needed for maintaining the desired pilot signal, of the digital sum value that the integration circuit  16 . 1  supplies. The squaring circuit  22 . 1  computes the energy of that deviation. 
     The computation in the PATH 0  system of the amount by which the spectral energy distribution of the I-NRZI modulation, when the generation thereof continues based on a “positive”-information channel word from the precoder  6 . 1 , deviates from the desired notch at frequency f 2  is done in the following way. A multiplier  24 . 1  multiplies the output of the precoder  6 . 1  by a sine-wave system function sinω 2 t of frequency f 2 ; an integration circuit  26 . 1  integrates the product from the multiplier  24 . 1 ; and the squaring circuit  28 . 1  squares the integration results from integration circuit  26 . 1  for application to the weighted summation network  52 . 1 . A multiplier  30 . 1  multiplies the output of the precoder  6 . 1  by a cosine-wave system function cosω 2 t of frequency f 2 ; an integration circuit  32 . 1  integrates the product from the multiplier  30 . 1 , and the squaring circuit  34 . 1  squares the integration results from integration circuit  32 . 1  for application to the weighted summation network  52 . 1 . (The phrase “system function” is used in digital electronics to refer to a function in the analog regime that is described on a sampled-data basis by digital samples.) 
     The computation in the PATH 0  system of the amount by which the spectral energy distribution of the I-NRZI modulation, when the generation thereof continues based on a “positive”-information channel word from the precoder  6 . 1 , deviates from the desired dip on either side of a peak at frequency f 1  is performed in the following way. A subtractor  36 . 1  subtracts a square wave of frequency f 1  supplied by the square wave generator  38  from the output signal of the precoder  6 . 1 . The square wave generator  38  and the subtractor  36 . 1  provide detection circuitry for detecting any deviation from the prescribed square wave of the “positive”-information serial-bit channel word that the precoder  6 . 1  supplies, as converted to arithmetic form by the code-to-arithmetic mapper  10 . 1 . A multiplier  40 . 1  multiplies the subtractor  36 . 1  difference output signal by a sine-wave system function sinω 1 t of frequency f 1 ; an integration circuit  42 . 1  integrates the product from the multiplier  40 . 1 ; and a squaring circuit  44 . 1  squares the integration results from the integration circuit  42 . 1  for application to the weighted summation network  52 . 1 . A multiplier  46 . 1  multiplies the subtractor  36 . 1  difference output signal by a cosine-wave system function cosω 1 t of frequency f 1 , an integration circuit  48 . 1  integrates the product from the multiplier  46 . 1 , and a squaring circuit  50 . 1  squares the integration results from the integration circuit  48 . 1  for application to the weighted summation network  52 . 1 . Filter circuitry PATH 1  determines the amount by which the spectral energy distribution of the I-NRZI modulation, when the generation thereof continues based on a “negative”-information channel word from the precoder  6 . 2 , deviates from the desired spectral energy distribution for a track recorded with the F 1  pattern having a peak at frequency f 1 , a dip on either side of frequency f 1  and a notch at frequency f 2 . A weighted summation circuit  52 . 2  in the filter circuitry PATH 1  combines with appropriate weighting the computed deviation from the desired notch at zero frequency and the desired peak at frequency f 1 , as furnished from a squaring circuit  22 . 2 , with the computed deviations from the other desired features as supplied from squaring circuits  28 . 2 ,  34 . 2 ,  44 . 2  and  50 . 2 . The weighted summation network  52 . 2  supplies, as the sum output signal therefrom, a second error signal e 2 . A comparator  54  compares the error signals e 1  and e 2  for generating the control signal CS, supplied to the selection control port of the selector  12  which selects the channel word having an error signal of a smaller value. 
     The computation in the PATH 1  system of the amount by which the spectral energy distribution of the I-NRZI modulation, when the generation thereof continues based on a “negative”-information channel word from the precoder  6 . 2 , deviates from the desired notch at zero frequency and the desired peak at frequency f 1  peak is done in the following way. An integration circuit  16 . 2  receives the current “negative”-information channel word from the precoder  6 . 2 , as converted to arithmetic form by the code-to-arithmetic mapper  10 . 2 , and integrates it with a prestored value. A subtractor  20 . 2  subtracts the output signal of the triangular wave generator  18  from the output of the integration circuit  16 . 2 , and a squaring circuit  22 . 2  for multiplies the resulting difference by itself; and the resulting square is supplied to the weighted summation network  52 . 2  to provide a component of the second error signal e 2 . The triangular wave generator  18  and the subtractor  20 . 2  provide detection circuitry for detecting any deviation from the prescribed digital sum needed for maintaining the desired pilot signal, of the digital sum value that the integration circuit  16 . 2  supplies. The squaring circuit  22 . 2  computes the energy of that deviation. 
     The computation in the PATH 1  system of the amount by which the spectral energy distribution of the I-NRZI modulation, when the generation thereof continues based on a “negative”-information channel word from the precoder  6 . 2 , deviates from the desired notch at frequency f 2  is done in the following way. A multiplier  24 . 2  multiplies the output of the precoder  6 . 2  by a sine-wave system function sinω 2 t of frequency f 2 ; an integration circuit  26 . 2  integrates the product from the multiplier  24 . 1 ; and the squaring circuit  28 . 2  squares the integration results from integration circuit  26 . 2  for application to the weighted summation network  52 . 2 . A multiplier  30 . 2  multiplies the output of the precoder  6 . 2  by a cosine-wave system function cosω 2 t of frequency f 2 ; an integration circuit  32 . 2  integrates the product from the multiplier  30 . 2 , and the squaring circuit  34 . 2  squares the integration results from integration circuit  32 . 2  for application to the weighted summation network  52 . 1 . 
     The computation in the PATH 1  system of the amount by which the spectral energy distribution of the I-NRZI modulation, when the generation thereof continues based on a “negative”-information channel word from the precoder  6 . 2 , deviates from the desired dip on either side of a peak at frequency f 1  is performed in the following way. A subtractor  36 . 2  subtracts a square wave of frequency f 1  supplied by the square wave generator  38  from the output signal of the precoder  6 . 2 . The square wave generator  38  and the subtractor  36 . 2  provide detection circuitry for detecting any deviation from the prescribed square wave of the “negative”-information serial-bit channel word that the precoder  6 . 2  supplies, as converted to arithmetic form by the code-to-arithmetic mapper  10 . 2 . A multiplier  40 . 2  multiplies the subtractor  36 . 2  difference output signal by a sine-wave system function sinω 1 t of frequency f 1 ; an integration circuit  42 . 2  integrates the product from the multiplier  40 . 2 ; and a squaring circuit  44 . 2  squares the integration results from the integration circuit  42 . 2  for application to the weighted summation network  52 . 2 . A multiplier  46 . 2  multiplies the subtractor  36 . 2  difference output signal by a cosine-wave system function cosω 1 t of frequency f 1 , an integration circuit  48 . 2  integrates the product from the multiplier  46 . 2 , and a squaring circuit  50 . 2  squares the integration results from the integration circuit  48 . 2  for application to the weighted summation network  52 . 2 . 
     The operation of the control signal generator  10  when generating the F 1  pattern has been described. When generating the F 2  pattern, the operation of the control signal generator  10  is modified by transposing f 1  and f 2 , thereby also transposing ω 1  and ω 2 . When generating the F 0  pattern, the operation of the control signal generator  10  is modified, disabling the triangular wave generator  18  and disabling the square wave generator  38 . Irrespective of whether the F 0 , F 1  or F 2  pattern is being generated, certain re-initialization procedures have to be followed subsequent to the decision being made as to whether to select a “positive”-information channel word provided by the precoder  6 . 1  or to select a “negative”-information channel word provided by the precoder  6 . 2  to determine the I-NRZI modulation to be recorded. These re-initialization procedures provide for continuity of coding and for enabling the control signal generator to establish a basis from which a decision can be made concerning which of the next pair of channel words is to be selected for recording. 
     In the latter regard, when the channel word that is to be recorded next has been determined, the contents of the integration circuits  16 . 1 ,  26 . 1 ,  32 . 1 ,  42 . 1  and  48 . 1  or the contents of the integration circuits  16 . 2 ,  26 . 2 ,  32 . 2 ,  42 . 2  and  48 . 2  have to be changed. If the newly selected channel word is of “negative”-information type, the contents of the integration circuits  16 . 1 ,  26 . 1 ,  32 . 1 ,  42 . 1  and  48 . 1  are changed to correspond to the contents of the integration circuits  16 . 2 ,  26 . 2 ,  32 . 2 ,  42 . 2  and  48 . 2 , respectively. If the newly selected channel word is of “positive”-information type the contents of the integration circuits  16 . 2 ,  26 . 2 ,  32 . 2 ,  42 . 2  and  48 . 2  are changed to correspond to the contents of the integration circuits  16 . 1 ,  26 . 1 ,  32 . 1 ,  42 . 1  and  48 . 1 , respectively. 
     As noted previously, when the channel word that is to be recorded next has been determined, preceding information from the “integrating feedback connection” of the one of the precoders  6 . 1  and  6 . 2  supplying the channel word which is selected for recording must be transferred into the “integrating feedback connection” of the other of the precoders  6 . 1  and  6 . 2 . If the channel word selected for being recorded next was supplied from the precoder  6 . 1 , the contents of the shift register in the integrating feedback connection of its exclusive-OR gate are transferred to corresponding positions in the shift register in the integrating feedback connection of the exclusive-OR gate in the precoder  6 . 2 . On the other hand, if the channel word selected for being recorded next was supplied from the precoder  6 . 2 , the contents of the shift register in the integrating feedback connection of its exclusive-OR gate are transferred to corresponding positions in the shift register in the integrating feedback connection of the exclusive-OR gate in the precoder  6 . 1 . 
     In practice, however, there is substantial time delay before this transfer can be completed in the prior-art digital signal recording apparatus described in U.S. Pat. No. 5,142,421, which delay arises in the multipliers, integration circuits, and squaring circuits in the control signal generator  10 . This delay which is particularly a problem when one attempts to digitize the control signal generator  10 , necessitates intermittently written buffer storage after the encoder  6 , as can be provided by the time compensator  8 , and necessitates intermittently read buffer storage before the encoder  6 , as can be provided by the parallel-to-serial converter  2 . The arrangements for this intermittent reading and writing of buffer storage are, in practice, difficult to arrange clocking for and can be avoided in accordance with the invention by performing precoding on a parallel-bit-word basis. 
     Referring to FIG. 6, an input port  101  for receiving serially supplied 8-parallel-bit words connects to the input port of a parallel-to-parallel (P/P) converter  102 . The P/P converter  102  converts each consecutive group of three serial 8-parallel-bit words supplied to its input port into three parallel 8-parallel-bit digital words, i.e., a 24-bit information word, and supplies the converted word in parallel-bit form from its output port  103 . 
     A signal inserting portion  104  affixes a single-bit digital word prefix to each 24-bit information word supplied in parallel-bit form from the output port  103  of the P/P converter  102 . The signal inserting portion comprises a “0” bit inserter  104 . 1  for affixing a “0” bit as prefix to the 24-bit information word, and a “1” bit inserter  104 . 2  for affixing a “1” bit as prefix to the 24-bit information word. 
     The thus-obtained 25-bit information words are supplied from output ports  105  and  107  to precoders  106 . 1  and  106 . 2 , respectively, of an encoding portion  106 . For the precoders  106 . 1  and  106 . 2 ,  2 T precoders are preferably used to convert a 25-bit information word into a 25-bit channel word. These  2 T precoders are suited for processing on a parallel-bit word basis and differ in their construction from those described in U.S. Pat. No. 5,142,421 suited for processing on a serial-bit word basis. The construction of these precoders  106 . 1  and  106 . 2 , each of which includes 25 exclusive-OR gates, will be described in detail further on in this specification with reference to FIGS. 8 and 10 of the drawing, in which the signal inserting portion  104  is included in the encoding portion  106 . Precoding still requires that bits that will be recorded later be determined based upon bits that will be recorded earlier. So time is required during preceding for ripple-through integration of the initialization bits and the successive bits used to form each channel word. However, the time required during precoding for ripple-through integration of these bits is only a fraction of the channel word interval. 
     The input ports of parallel-to-serial (P/S) converters  108 . 1  and  108 . 2  of a first signal converter  108  respectively connect from output ports  109  and  111  of the precoders  106 . 1  and  106 . 2 ; and the output ports of converters  108 . 1  and  108 . 2  respectively connect to input ports of delays  114 . 1  and  114 . 2  of a time compensator  114 . Each of the converters  108 . 1  and  108 . 2  converts each 25-parallel-bit channel word supplied thereto into a 25-serial-bit channel word supplied at the bit rate associated with the I-NRZI modulation recorded on the magnetic recording medium. 
     P/S converters  110 . 1  and  110 . 2  of a second signal converter  110  convert to serial-bit form the odd-numbered bit-places of each channel word (hereinafter referred to as an “odd channel” word) from the 25-bit channel words supplied in parallel from the precoders  106 . 1  and  106 . 2 . P/S converters  112 . 1  and  112 . 2  of a third signal converter  112  convert to serial-bit form the even-numbered bit-places of each channel word (hereinafter referred to as an “even channel” word) from the 25-bit channel words supplied in parallel from the precoders  106 . 1  and  106 . 2 , respectively. 
     Fixed delays created by delay elements  114 . 1  and  114 . 2  of time compensator  114  compensate for the time taken by a control signal generator  116  to generate a control signal indicating to a selector  118  which of the channel words respectively generated by the precoders  106 . 1  and  106 . 2  and delayed by the delay elements  114 . 1  and  114 . 2  to select to a recording portion  120 . 
     The control signal generator  116  generates first, second and third control signals CS 1 , CS 2  and CS 3  on the basis of the channel word signals supplied respectively from the respective output ports  117 ,  119 ,  121  and  123  of the P/S converters  110 . 1 ,  110 . 2 ,  112 . 1  and  112 . 2 . The circuitry in the control signal generator  116  that decides which of the channel words generated by the precoders  106 . 1  and  106 . 2  is to be recorded, processes the odd-channel word supplied from the P/S converter  110 . 1  and the even-channel word supplied from the P/S converter  112 . 1  in parallel, and this circuitry also processes the odd-channel word supplied from the P/S converter  110 . 2  and the even-channel word supplied from the P/S converter  112 . 2  in parallel. These parallel processing procedures halve the time required to complete the decision procedure, the computations for which are clocked at the same bit rate as the I-NRZI signal that is to be recorded. Accordingly, the computations can be completed in a little over half the time interval between serial-word channel word clocks that occur at one-twenty-fifth the bit rate of the I-NRZI signal that is to be recorded. The time for these computations combined with the time for ripple-through integration in the precoders  106 . 1  and  106 . 2  of the encoding portion  106  is sufficiently less than the time interval between channel word clocks, to afford plenty of time to re-initialize integrators within the control signal generator  116  and to set up initialization for ripple-through integration that is to take place when the next serial-word is clocked into the precoders  106 . 1  and  106 . 2 . The first and second control signals CS 1  and CS 2  that the control signal generator  116  supplies via its output ports  125  and  127  are applied to the respective control ports of the precoders  106 . 1  and  106 . 2 . The third control signal CS 3  the control signal generator  116  supplies via its output port  127  is applied to the selection control port of the selector  118 . 
     In accordance with the third control signal CS 3 , the selector  118  selects a value closer to an intended frequency characteristic between the 25-serial-bit “positive”-information channel word supplied by the P/S converter  108 . 1  and the 25-serial-bit “odd”-information channel word supplied by the P/S converter  108 . 2 , and transmits the selected word to the recording portion  120 . 
     Reductions can be made in the FIG. 6 digital signal recording apparatus. Corresponding bit places of the even channel words supplied in parallel from the precoders  106 . 1  and  106 . 2  are identical if they are of  2 T type and single-bit prefixes are used, so one of the P/S converters  112 . 1  and  112 . 2  can be dispensed with, and the signal supplied from its output port to the control signal generator  116  can be supplied instead from the output port of the remaining one of the converters  112 . 1  and  112 . 2 . If the precoders  106 . 1  and  106 . 2  are of  2 T type and single-bit prefixes are used, corresponding bit places of the odd channel words they supply in parallel are bit complements of each other, so one of the P/S converters  110 . 1  and  110 . 2  can be dispensed with, and the signal supplied from its output port to the control signal generator  116  can be supplied instead by bit-complementing the signal from the output port of the remaining one of the converters  110 . 1  and  110 . 2 . 
     FIG. 7 is a block diagram of another embodiment of the digital signal recording apparatus of the present invention. In the drawing, the same numerals designate the same components as the apparatus of FIG.  6 . Accordingly, configuration and operation that are the same will not be described again. Referring to FIG. 7, output ports  117 ′,  119 ′,  121 ′ and  123 ′ of the first signal converter  108  are coupled directly to the input ports of a modified control signal generator  116 ′, such that the second and third signal converters  110  and  112  of FIG. 6 are eliminated from the circuit. 
     In the operation of FIG. 7, responsive to the 25-parallel-bit “positive”-information channel word supplied from the precoder  106 . 1 , a P/S converter  108 . 3  within the first signal converter  108  supplies first through thirteenth bits of the channel word (hereinafter referred to as the “leading bit group”) via output port  117 ′ to the control signal generator  116 ′. At the same time the P/S converter  108 . 3  supplies the first through twelfth of these bits, it also supplies fourteenth through twenty-fifth bits of the channel word (hereinafter referred to as the “trailing bit group”) via output port  121 ′ to the control signal generator  116 ′. 
     Responsive to the 25-parallel-bit “negative”-information channel word supplied from precoder  106 . 2 , a P/S converter  108 . 4  within the first signal converter  108  supplies first through thirteenth bits of the channel word (hereinafter referred to as the “leading bit group”) via output port  119 ′ to the control signal generator  116 ′. During the same time the P/S converter  108 . 4  also supplies fourteenth through twenty-fifth bits of the channel word (hereinafter referred to as the “trailing bit group”) via the output port  123 ′ to the control signal generator  116 ′. 
     The modified control signal generator  116 ′ performs the same general calculations as the control signal generator  116 , but in somewhat different order, requiring modifications of the FIG. 4 filter circuitry in regard to the triangular wave generator  18 , the square wave generator  38  and the sine and cosine signal generators. These modifications are readily made by one of ordinary skill in the art of digital system design. This is particularly so where these generators are implemented using read-only memory (ROM), since the order of the sequential reading of the samples of each of the various system functions is readily permuted. 
     FIG. 8 is a detailed block diagram of the “0” bit inserter  104 . 1 , the precoder  106 . 1  and the P/S converters  108 . 1 ,  110 . 1  and  112 . 1 , each of which is shown in FIG.  6 . Referring to FIG. 8, the “0” bit inserter  104 . 1  is made up of  25  latches  104 .a through  104 .y. A “0” bit is applied to the latch  104 .a which stores the most significant bit, according to a system clock (CLOCK  1 ) and a load command signal LOAD. The remaining latches  104 .b through  104 .y receive the 24-bit information word supplied in parallel from the output port  103  of the P/P converter  102 . 
     As shown in FIG. 9, which is a detailed circuit diagram of the “0” bit inserter  104 . 1 , each of the 25 latches is made up of one D flip-flop, two AND gates and one OR gate. In the operation of the inserting portion  104 . 1 , when the LOAD command signal is a logic high, a “0” bit applied to the data port of the latch  104 .a and the 24-bit information word supplied from the P/P converter  102  are latched and supplied from the Q outputs of the respective D flip-flops. When the LOAD command signal is a logic low, the latches maintain the output of each D flip-flop. 
     The first input ports of XOR gates  106 .a through  106 .y of the precoder  106 . 1  shown in FIG. 8 are respectively coupled to the respective output ports of the latches  104 .a through  104 .y of the “0” bit inserter  104 . 1 . The second inputs of the XOR gates  106 .a and  106 .b are tied to the respective outputs of the latches  106 . 3  and  106 . 4 . The respective outputs of the XOR gates  106 .a through  106 .w connect to the second inputs of the XOR gates  106 .c through  106 .y. The outputs of the XOR gates  106 .x and  106 .y are coupled to the respective inputs of the latches  106 . 3  and  106 . 4 . 
     The operation of precoder  106  will be explained below. 
     The second least significant bit from the preceding channel word and the MSB (here, the inserted “0” bit) of the current 25-bit channel word are supplied to the XOR gate  106 .a. The least significant bit (LSB) from the preceding channel word and the second MSB bit (here, the first bit of input data) of the current 25-bit channel word are supplied to the XOR gate  106 .b. The output of the XOR gate  106 .a and the second bit of the input data are supplied to the XOR gate  106 .c. The output of the XOR gate  106 .b and the third bit of the input data are supplied to the XOR gate  106 .d. 
     The XOR gates  106 .e through  106 .y precode the remaining data of the 25-bit channel word in similar manner. The outputs of the XOR gates  106 .a through  106 .y are the 25-bit channel word (precoded data) supplied in parallel from the precoder  106 . 1 . 
     FIG. 10 is a detailed circuit diagram of the latches  106 . 3  and  106 . 4  of the precoder  106 . 1 . Referring to FIG. 10, when the LOAD signal is a logic high, output signal  24  of the XOR gate  106 .x supplied to the data port of a D flip-flop D 2  via gates G 8  and G 9  is applied as the second LSB  24 ′ of the preceding channel word, to the second input of the XOR gate  106 .a of FIG. 8 according to the system clock (CLOCK  1 ). Simultaneously, output signal  25  of the XOR gate  106 .y applied to the data port of a D flip-flop D 1  via gates G 2 , G 3 , G 5  and G 6  is supplied as the LSB  25 ′ of the preceding channel word, to the second input port of the XOR gate  106 .b of FIG. 8 according to the system clock signal (CLOCK  1 ). While the LOAD command signal is low (and until it goes high), the Q outputs of the D flip-flops D 1  and D 2  are maintained. 
     Since the output of the D flip-flop D 1  is subject to the influence of the first control signal CS 1  supplied from the first control signal output port  125  of the control signal generator  116  shown in FIG. 6, if first control signal CS 1  is high, the output  25  of the XOR gate  106 .y is supplied to the gate G 2  without change. If the first control signal CS 1  is low, the output of the XOR gate  106 .y is complemented. 
     For instance, when the output  25  of the XOR gate  106 .y is a logic high and the first control signal CS 1  is a logic low, the output of the D flip-flop D 1  is low. If the first control signal CS 1  and the output  25  are both high, the output of the D flip-flop D 1  is high. 
     If the first control signal CS 1  is a logic high, which indicates that the “positive”-information channel word is selected, the initial value of the latch  106 . 3  of the precoder  106 . 1  stays unchanged. If the control signal CS 1  is a logic low, which indicates that the “negative”-information channel word is selected, the initial value of the latch  106 . 3  of the precoder  106 . 1  is complemented. 
     P/S converter  108 . 1  of FIG. 8 receives the respective outputs of the XOR gates  106 .a through  106 .y in parallel according to the system clock and LOAD command signal, thereby supplying the received outputs as a serial 25-bit channel word. FIG. 8 shows the P/S converter  108 . 1  is composed of 25 latches  108 .a through  108 .y, which FIG. 11 shows in detail. FIG. 11 shows each latch being made up of two AND gates, an OR gate and a D flip-flop. 
     When the LOAD command signal is a logic high, the D flip-flops each receive the output of a corresponding XOR gate of the precoder  106 . 1  and supply it as the input of the first AND gate of the latch of the next higher bit. If the LOAD command signal is a logic low, each D flip-flop holds its Q output until the LOAD command signal goes high. As the final output, a serial 25-bit channel word is supplied from the output port  113 . 
     The P/S converter  108 . 3  of FIG. 7 has the same configuration as that of the P/S converter  108 . 1  of FIG.  11 . However, the difference is that output port  117 ′ of the latch  108 .a and the output port  121 ′ of the latch  108 .n are coupled to the control signal generator  116 . 
     The P/S converter  110 . 1  of FIG. 8 is composed of thirteen latches  110 .a,  110 .c, . . . , and  110 .y. Their configuration is the same as that of the respective latches of the P/S converter  108 . 1  shown in FIG.  11 . Responsive to the LOAD command signal and clock signal simultaneously occurring, odd channel words are selected from the 25-bit channel word (supplied in parallel from the precoder  106 . 1 ) to be loaded in parallel into these thirteen latches  110 .a,  110 .c, . . . , and  110 .y, so that a 13-bit odd channel word is supplied serially from the output port  117  of the latch  110 .a. 
     The P/S converter  112 . 1  of FIG. 8 has 12 latches  112 .b,  112 .d, . . . , and  112 .x. Their configuration is the same as that of the latches of the P/S converter  108 . 1  shown in FIG.  11 . Responsive to the LOAD command signal and clock signal simultaneously occurring, even channel words are selected from the 25-bit channel word (supplied in parallel from the precoder  106 . 1 ) to be loaded in parallel into these 12 latches  112 .b,  112 .d, . . . , and  112 .x, so that a 12-bit even channel word is supplied serially from the output port  121  of the latch  112 .a. 
     FIG. 12 is a block diagram of the control signal generator  116  shown in FIG. 6, which includes a PATH 0  unit  116 . 1 , a PATH 1  unit  116 . 2 , a detector  116 . 3 , and code-to-arithmetic mappers  116 . 4 - 116 . 7 . The code-to-arithmetic mapper  116 . 4  converts the ONEs and ZEROs supplied from the output port  117  of the P/S converter  110 . 1  of FIG. 6 to arithmetic descriptions of NRZI modulation that switches between negative and positive arithmetic values of similar amplitude and is unaccompanied by a direct term, which arithmetic descriptions are supplied from the output port  117 ′ of the code-to-arithmetic mapper  116 . 4 . A code-to-arithmetic mapper  116 . 5  converts the ONEs and ZEROs supplied from the output port  121  of the P/S converter  110 . 1  of FIG. 6 to arithmetic descriptions of NRZI modulation that switches between negative and positive arithmetic values of similar amplitude and is unaccompanied by a direct term, which arithmetic descriptions are supplied from the output port  121 ′ of the code-to-arithmetic mapper  116 . 5 . A code-to-arithmetic mapper  116 . 6  converts the ONEs and ZEROs supplied from the output port  119  of the P/S converter  110 . 2  of FIG. 6 to arithmetic descriptions of NRZI modulation that switches between negative and positive arithmetic values of similar amplitude and is unaccompanied by a direct term, which arithmetic descriptions are supplied from the output port  119 ′ of the code-to-arithmetic mapper  116 . 6 . A code-to-arithmetic mapper  116 . 7  converts the ONEs and ZEROs supplied from the output port  123  of the P/S converter  110 . 2  of FIG. 6 to arithmetic descriptions of NRZI modulation that switches between negative and positive arithmetic values of similar amplitude and is unaccompanied by a direct term, which arithmetic descriptions are supplied from the output port  123 ′ of the code-to-arithmetic mapper  116 . 6 . 
     The first and second input ports of a PATH 0  unit  116 . 1  connect to the respective output ports  117 ′ and  121 ′ of the code-to-arithmetic mappers  116 . 4  and  116 . 5 . Preset signal output port  137  of a PATH 1  unit  116 . 2  is connected to the preset input port of the PATH 0  unit  116 . 1 . The output port of the PATH 0  unit  116 . 1  for supplying error signal e 1  is coupled to the first input port of the detector  116 . 3 . The first and second input ports of PATH 1  unit  116 . 2  connect to the respective output ports  119 ′ and  123 ′ of the code-to-arithmetic mappers  116 . 6  and  116 . 7 . Preset signal output port  135  of the PATH 0  unit  116 . 1  is connected to the preset input of the PATH 1  unit  116 . 2 . The output port of the PATH 1  unit  116 . 2  for supplying error signal e 2  is coupled to the second input port of the detector  116 . 3 . The first and second control signal output ports  125  and  127  of the detector  116 . 3  are connected to the respective control ports of the precoders  106 . 1  and  106 . 2  of FIG.  6  and to the respective control ports of units  116 . 1  and  116 . 2 . Third control signal output port  129  is coupled to the selection control port of the selector  118 . 
     FIG. 13 is a detailed circuit diagram of the PATH 0  unit  116 . 1  of the control signal generator shown in FIG.  12 . The first and second input ports of the PATH 0  unit  116 . 1  connect to respective output ports  117 ′ and  121 ′ of the code-to-arithmetic mappers  116 . 4  and  116 . 5  of FIG. 12 to receive two&#39;s complement numbers descriptive of I-NRZI modulation that are used as input signal by arithmetic elements  122 ,  124 ,  134 ,  138 ,  146 ,  150 ,  158  and  174 . The unit  116 . 1  is composed of the integration circuits  122 ,  124  through a squaring circuit  132  for forming a pilot signal at an intended frequency (here, f 1 ) on the frequency spectrum of the 25-bit serial data stream while at the same time forming a notch at zero frequency, the multipliers  134 ,  138  through a squaring circuit  156  for forming a notch at an intended frequency (here, f 2 ), the subtractors  158 ,  174  through a squaring circuit  188  for forming dips on the skirts of the pilot signal (f 1 ), and a weighted summation network  190  for summing the outputs of squaring circuits  132 ,  144 ,  156 ,  172  and  188 , thereby generating error signal e 1 . 
     The odd channel word input from the output port  117 ′ and the even channel word input from the output port  121 ′ are added to a value (the digital sum value of the preceding 25-bit channel word) prestored in respective integration circuits  122  and  124 . The respective outputs of the integration circuits  122  and  124  are summed in an adder  126  and then supplied to the first input port of the subtractor  130 . 
     A triangular wave generator  128  is made up of a ROM and generates a triangular wave signal corresponding to the digital sum value (DSV) of the serial data stream of channel words being descriptive of a prescribed frequency (here, f 1 ), corresponding to the fundamental frequency component of the triangular wave signal. If the signal generated from the ROM is a triangular wave of frequency f 1  (for instance,  1 / 90 T) as shown in FIG. 14A, 8-bit data (for instance,  90 A through  90 L) is stored using 5-bit addresses which are indicative of values zero through sixteen in the ROM table shown in FIG.  14 B. The subtractor  130  subtracts the output of the triangular wave generator  128  from the output of the adder  126 . The difference value is squared in the squaring circuit  132  and applied to the weighted summation network  190 . The triangular wave generator  128  and the subtractor  130  provide detection circuitry for detecting any deviation from the prescribed digital sum needed for maintaining the desired pilot signal, of the digital sum value that the adder  126  supplies; and the squaring circuit  132  computes the energy of that deviation. These computations are to implement a notch being formed at f=0 Hz (in other words, the DC component) and a pilot signal being formed at frequency f 1 . 
     Computations are also made to implement the introduction of a notch at frequency f 2 (ω 2 /2π) by generating summand input signals for application to the weighted summation network  190  whenever there is energy at the frequency in the spectrum of the “positive”-information channel word supplied by the precoder  106 . 1 . This is done as follows. 
     A multiplier  134  multiplies the odd channel words by odd sine signal o_sinω 2 t, and the resulting product is integrated in an integration circuit  136 . A multiplier  138  multiplies the even channel words by even sine signal e_sinω 2 t, and the resulting product is integrated in an integration circuit  140 . The integration results from the integration circuits  136  and  140  are added in an adder  142 . The resulting sum is squared in the squaring circuit  144 , and the resulting square is applied to the weighted summation network  190 . 
     A multiplier  146  multiplies the odd channel words by odd cosine signal o_cosω 2 t, and the resulting product is integrated in an integration circuit  148 . The even channel words and even cosine signal e_cosω 2 t are multiplied together in a multiplier  150 , and the resulting product is integrated in an integration circuit  152 . An adder  154  sums the integration results from the integration circuits  148  and  152 . The summed value is squared by the squaring circuit  156  and the resulting square is supplied as a summand to the weighted summation network  190 . 
     A ROM (not shown) generates a sine signal input for application to the multipliers  134  and  138 . The sine table stored in the ROM is divided into an odd-sample sine table and an even-sample sine table. If the waveform of the sine signal is, for instance,  1 / 60 T for frequency f 2 , as shown in FIG. 15A, one period of the sine signal is divided into sixty addresses, and data corresponding to the amplitude of a sampled sine signal is stored in each address of the sine table. Data corresponding to the odd addresses of the sampled sine signal is stored in the odd-sample sine table. The even-sample sine table stores data corresponding to the even addresses of the sampled sine signal. As shown in FIG. 15B, the points corresponding to bits (indicated by dots) become alternately odd addresses or even addresses of the sine signal sampled by the period of 25-bit channel word. In the drawing, the characters EB (extra bit) indicate where a “0” bit is inserted, that is, the MSB. Similarly, the cosine signal supplied to the multipliers  146  and  150  may be generated by a ROM having an odd-sample cosine table and an even-sample cosine table. When the sine signal and cosine signal are designed to be generated by a single ROM, an address shifted by 45° with respect to the sine signal is applied and a corresponding value (the cosine) is read out. 
     A dip is also introduced in portions of the frequency spectrum flanking the frequency f 1 =(ω 1 /2π) by generating summand input signals for application to the weighted summation network  190  whenever there is energy in those portions of the frequency spectrum of the “positive”-information channel word supplied by the precoder  106 . 1 . This is done as follows. 
     A subtractor  158  subtracts, from the odd channel words, the odd samples of a sampled square wave signal (FIG. 15C) generated by a square wave generator  160 . The square wave generator  160  and the subtractor  158  provide detection circuitry for detecting any deviation from the prescribed square wave of the “positive”-information serial-bit odd channel word that the P/S converter  110 . 1  supplies, as converted to arithmetic form by the code-to-arithmetic mapper  116 . 4 . A multiplier  162  multiplies the output of the subtractor  158  by odd sine signal o_sinω 1 t, and the resulting product is integrated in an integration circuit  164 . A multiplier  166  multiplies the output of the subtractor  158  by odd cosine signal o_cosω 1 t, and the resulting product is integrated in an integration circuit  168 . 
     A subtractor  174  subtracts, from the even channel words, even samples of a sampled square wave signal generated by the square wave generator  176 . The square wave generator  176  and the subtractor  174  provide detection circuitry for detecting any deviation from the prescribed square wave of the “positive”-information serial-bit odd channel word that the P/S converter  112 . 1  supplies, as converted to arithmetic form by the code-to-arithmetic mapper  116 . 5 . A multiplier  178  multiplies the output of the subtractor  174  by even-sample sine signal o_sinω 1 t, and the resulting product is integrated in an integration circuit  180 . A multiplier  182  multiplies the output of the subtractor  174  by even-sample cosine signal o_cosω 1 t, and the resulting product is integrated in an integration circuit  184 . 
     An adder  170  sums the respective outputs of the integration circuits  164  and  180 ; the resulting sum is squared by the squaring circuit  172 ; and the squared result is applied to the weighted summation network  190 . An adder  186  sums the respective outputs of the integration circuits  168  and  184 ; the resulting sum is squared by the squaring circuit  188 ; and the squared result is supplied to the weighted summation network  190 . Then, the weighted summation network  190  sums the outputs of the squaring circuits  132 ,  144 ,  156 ,  172  and  188 , thereby generating error signal e 1 . 
     The operation shown in FIG. 13 is similarly performed in PATH 1  unit  116 . 2  of FIG.  12 . The difference is that the control signal input to the respective integration circuits (not shown) of unit  116 . 2  is second control signal CS 2 , and that error signal e 2  is generated from a weighted summation network (not shown) of unit  116 . 2 . When the precoders  106 . 1  and  106 . 2  are of  2 T type, certain of the computations carried out in PATH 0  and in PATH 1  before integration procedures are similar in nature, permitting some sharing of hardware, if desired. The error signal e 1  is indicative of how much the DSV in the serial data stream formed by next selecting the “positive”-information word deviates from a prescribed DSV; and the error signal e 2  is indicative of how much the DSV in the serial data stream formed by next selecting the “negative”-information word deviates from that prescribed DSV. If the error signal e 1  is smaller than the error signal e 2 , the “positive”-information word from the precoder  106 . 1  will be selected for recording. If the error signal e 2  is smaller than the error signal e 1 , the “negative”-information word from the precoder  106 . 2  will be selected for recording. If the error signals e 1  and e 2  are alike, it is preferable to record the “positive”-information word from the precoder  106 . 1 . 
     The detector  116 . 3  of FIG. 12 includes a comparator which selects the smaller value between error signals e 1  and e 2  and supplies the third control signal CS 3 . The comparator is typically formed as a two&#39;s complement subtractor receptive of error signals e 1  and e 2  with “0” bit sign extensions as minuend and subtrahend, the sign bit of the resulting difference being used as the third control signal CS 3 . The third control signal CS 3  determines which of the first and second control signals CS 1  and CS 2  will be generated at a time close to the end of the channel word interval. 
     According to first and second control signals CS 1  and CS 2  generated from the detector  116 . 3  of FIG. 12, that is, when first control signal CS 1  is high and second control signal CS 2  is low, PATH 0  having error signal e 1  is selected so that the values of the respective integration circuits of PATH 1  are replaced with the values stored in the respective integration circuits  122 ,  124 ,  136 ,  140 ,  148 ,  152 ,  164 ,  168 ,  182 , and  184  corresponding to PATH 0  shown in FIG. 13 via preset output port  131 . 
     FIGS. 16A-16G are operation waveform diagrams of blocks shown in FIG.  6 . 
     FIG. 16A illustrates the output waveform of the P/S converter  108 . 1  of the first converter  108  for converting the “positive”-information 25-parallel-bit channel word supplied from the encoding portion  106  into a 25-serial-bit channel word according to the system clock (CLOCK  1 ) shown in FIG.  16 D. FIG. 16B illustrates the output waveform of the P/S converter  110 . 1  of the second converter  110  for receiving the “positive”-information 25-parallel-bit channel word from the encoding portion  106  and serially supplying only the odd channel words selected therefrom, as clocked in accordance with the system clock (FIG.  16 D). FIG. 16C illustrates the output waveform of the P/S converter  112 . 1  of the third converter  112  for receiving the “positive”-information 25-parallel-bit channel word from the encoding portion  106  and serially supplying only the even channel words selected therefrom, as clocked in accordance with the system clock 
     FIGS. 16E,  16 F and  16 G illustrate first, second and third control signals CS 1 , CS 2  and CS 3  generated by the control signal generator  116 . 
     The first and second control signals CS 1  and CS 2  are alternately high at the ends of cycles of 25 bits length. The first and second control signals CS 1  and CS 2  are respectively supplied to the first precoder  106 . 1  and to the second precoder  106 . 2 . The third control signal CS 3  is supplied to the selector  118 . If the third control signal CS 3  is high, the selector  118  selects the output of the P/S converter  108 . 1  as delayed by the delay  114 . 1  throughout the ensuing cycle of 25 bits length. If the third control signal CS 3  is low, the selector  118  selects the output of the P/S converter  108 . 2  as delayed by the delay  114 . 2  throughout the ensuing cycle of 25 bits length. 
     Therefore, if the data is time-share-multiplexed into the odd channel words and even channel words shown in FIGS. 16B and 16C, although delayed by the integration circuits, multipliers and squaring circuits of the control signal generator shown in FIG. 13, a reduction of at least twelve system clocks is provided for in the time required to compute a control signal, compared to the period of 25 system clocks per channel word. If the data is time-share-multiplexed into leading and trailing bit groups, a similar reduction is possible in the time required to compute a control signal. This enables a control signal to be generated in real time for selecting one output, that is, the one having the intended spectral characteristics, from between those supplied from the P/S converters  108 . 1  and  108 . 2 . 
     FIG. 17 is another detailed circuit diagram of PATH 0  shown in FIG. 12, showing reductions that can be made in the FIG. 13 PATH 0  circuit. The two integration circuits  122  and  124  and the single adder  126  surrounded by a dashed line in FIG. 13 are replaced in FIG. 17 by a simpler, equivalent circuit made up of a single adder  192  and a single integration circuit  194 . The two integration circuits  136  and  140  and the single adder  142  surrounded by a dashed line in FIG. 13 are replaced in FIG. 17 by a simpler, equivalent circuit made up of a single adder  206  and a single integration circuit  208 . The two integration circuits  148  and  152  and the single adder  154  surrounded by a dashed line in FIG. 13 are replaced in FIG. 17 by a simpler, equivalent circuit made up of a single adder  216  and a single integration circuit  218 . The two integration circuits  164  and  180  and the single adder  170  surrounded by a dashed line in FIG. 13 are replaced in FIG. 17 by a simpler, equivalent circuit made up of a single adder  230  and a single integration circuit  232 . And the two integration circuits  168  and  184  and the single adder  186  surrounded by the same dashed line in FIG. 13 are replaced in FIG. 17 by a simpler, equivalent circuit made up of a single adder  244  and a single integration circuit  246 . When the precoders  106 . 1  and  106 . 2  are of  2 T type, certain of the computations carried out in PATH 0  and in PATH 1  before integration procedures are similar in nature, permitting some sharing of hardware, if desired. 
     FIG. 18 shows another digital signal recording apparatus embodying the invention in which the parallel-bit words serially supplied from the precoders are converted to serial-bit format with a bit rate that is a multiple of the bit rate used during digital recording. Components that are the same as those used in FIG. 6 are numbered with the same numerals, and description of their operation will not be repeated. 
     The configuration of FIG. 18 is the same as that of FIG. 6, except for a second converter  310  for converting the 25-bit channel word supplied in parallel from the encoding portion  106  into a serial 25-bit channel word according to a second clock (CLOCK  2 ) of twice the frequency of the system clock signal (CLOCK  1 ). The second converter  310  replaces both the second converter  110  for converting the odd channel words from the 25-parallel-bit channel word supplied from the encoding portion  106  of FIG. 6 into a serial-bit channel word and the third converter  112  for converting the even channel words from the 25-parallel-bit channel word supplied from the encoding portion  106  into a serial-bit channel word. 
     The operation of FIG. 18 will be explained with reference to FIGS. 19A through 19D. 
     In FIG. 18, the detailed configuration and operation of the P/P converter  102 , the signal inserting portion  104 , the encoding portion  106 , and the first converter  108  are the same as those in FIGS. 8 through 11. 
     FIG. 19A illustrates the output waveform of the P/S converter  108 . 1  of the first P/S converter  108 , which converts the 25-parallel-bit “positive”-information channel word (as supplied from the precoder  106 . 1 ) into a 25-serial-bit “positive”-information channel word. 
     FIG. 19B shows the first clock signal (CLOCK  1 ), in accordance with which the serial-bit signals from the first converter  108  are clocked. 
     FIG. 19C illustrates the output waveform of the P/S converter  310 . 1  of the second converter  310  for converting the 25-parallel-bit “positive”-information channel word (as supplied from the precoder  106 . 1 ) into a 25-serial-bit channel word supplied at a bit rate twice as high as the 25-serial-bit channel word supplied from the P/S converter  108 . 1 . 
     FIG. 19D shows the second clock signal (CLOCK  2 ), in accordance with which the serial-bit signals from the second converter  310  are clocked. 
     In the FIG. 18 digital signal recording apparatus, the control signal generator  116 ″ receives the output of the second converter  310  which is time-compressed twofold in accordance with the second clock signal and thereby corresponds to half the original period of the 25-parallel-bit channel word. The comparison between the respective frequency components of the time-compressed “positive”-information 25-serial-bit channel words and of the time-compressed “negative”-information 25-serial-bit channel words supplied in parallel is carried out well within one 25-parallel-bit channel-word interval, despite delay introduced into the computations by the integration circuits, multipliers and squaring circuits of the control signal generator  116 ″ similar to those shown in FIG. 13 or  17 . Accordingly, a control signal for selecting a 25-parallel-bit channel word for an intended channel can be generated without having to depart from pipeline processing of channel words. Twofold time compression is generally sufficient and is preferred, because of the ease with which the clock signals with rates in 2:1 ratio can be generated using simple counter circuitry, and because doubling of the clocking rate does not tend to require an excessively high clock rate. 
     Other alternative embodiments of the invention, in addition to those thusfar described, will be apparent to one skilled in the art of digital tape recorder design and acquainted with the foregoing specification; and such alternative embodiments are intended to be considered as being within the scope of the claims appended to this specification. By way of specific example, the time compensator  114  after the first converter  108  used to delay the output signals from the precoders  106 . 1  and  106 . 2  as applied to the selector  118  not only can be fixed delay owing to the invention, but in certain designs of the sort shown in FIG. 6 time compensation can be obtained at least in part by delaying the latching of channel words from the  2 T precoders  106 . 1  and  106 . 2  into the P/S converters  108 . 1  and  108 . 2 . By way of further specific example, in other embodiments of the invention the delays of the output signals from the precoders  106 . 1  and  106 . 2  as applied to the selector  118  are introduced before the first converter  108  (e. g., by respective word latches), rather than being provided after the first converter  108 . In yet other embodiments of the invention, the selection between the output signals from the precoders  106 . 1  and  106 . 2  is performed while the signals are still in 25-parallel-bit format, and conversion to serial-bit format for recording is deferred until after the selection between channel words is completed. 
     The triangular wave generator  128  of FIG. 13 can be replaced by a triangular wave generator generating a triangular wave complementary to that generated by the generator  128 , and the subtractor  130  replaced by an adder, without changing operation. The square wave generators  160  and  176  of FIG. 13 can be replaced by square wave generators generating square waves complementary to those generated by the generators  160  and  176 , and the subtractors  158  and  174  replaced by respective adders, without changing operation. Analogous modifications can be made in the portions of the control signal generators shown in FIGS. 4 and 17. 
     Methods of estimating the energies of deviations from their absolute values, rather than squaring the deviations, are known to digital designers, and circuitry using such methods are equivalents of the squaring circuitry shown in FIGS. 13 and 17. Embodiments of the invention wherein the precoders  106 . 1  and  106 . 2  are of an aT type where a is three or is a still higher integer are also envisioned.