Abstract:
Pre-emphasis circuitry and methods for signal transmission provide multiple levels of output signal amplification over one or more baud periods after an input signal transition. The multiple, gradually decreasing levels of output signal amplification reduce power consumption and better approximate the desired signal response.

Description:
BACKGROUND OF THE INVENTION 
   This invention relates to signal transmission circuitry and methods. More particularly, this invention relates to pre-emphasis of data signals to improve signal transmission quality. 
   Signals transmitted at high frequencies and low voltages are particularly susceptible to signal losses over long traces. Traces are signal transmission paths through signal wiring, integrated circuit bus structures, PCBs (printed circuit boards), etc. Signal losses can be caused by, for example, attenuation, which is a decrease in the power of a signal, crosstalk, which is an adverse effect caused by signal transmission on an adjacent trace, and intersymbol interference, which is an adverse effect caused by residual noise from a previously transmitted signal. These losses can adversely affect the speed and accuracy at which transmitted data is received. For example, a logical 1 data signal may be incorrectly received as a logical 0 data signal and vice versa. An entire system can therefore be adversely affected by such signal transmission losses. 
   To compensate for such losses, signals may be “pre-emphasized.” Pre-emphasis is extra power (usually in the form of extra voltage, but extra current may be an equivalent) briefly applied to a transmitted signal immediately adjacent each signal transition (i.e., a signal changing from a logical 0 to logical 1 and vice versa). Pre-emphasis helps more quickly change the state of the medium transmitting the signal to receiver circuitry, and helps the receiver circuitry respond more rapidly to the change in state of the transmitted signal. 
   Pre-emphasis is becoming increasingly important as communication protocols and standards call for lower and lower signaling voltages (or currents) and increased signaling speeds. For example, very low signaling voltages are being specified for low voltage differential signaling (“LVDS”) and current mode logic (“CML”) communication protocols. A typical CML protocol may have a voltage swing of only 0.4 volts. At the same time, such a protocol may specify data transmission in the gigabit (i.e., billion bits) per second range. At such low voltages and high data rates, transmission line losses become a serious impediment to accurate and error-free reception of transmitted data. 
   Known pre-emphasis circuitries typically amplify a data signal at a constant amplitude level for the full duration of a baud period. A baud period can be generally thought of as the minimum amount of time between signal transitions. Such pre-emphasis, while improving signal transmission quality somewhat, does not adequately approximate the desired pulse shape of the transmitted signal. Thus, transmitted signals still lack the robustness desired for long traces and are accordingly still subject to transmission losses from, for example, crosstalk and residual noise. Moreover, because such known pre-emphasis is applied at a constant amplitude for the entire baud period, it results in high power consumption. Accordingly, known pre-emphasis circuitries can benefit from improvement. 
   SUMMARY OF THE INVENTION 
   In accordance with the invention, pre-emphasis circuitries and methods are provided that pre-emphasize signal transitions via a series of amplitude levels or steps rather than via a single amplitude level as is known. Such a series of levels better approximates the desired signal pulse shape and thus improves the speed and accuracy at which data can be received, particularly over long traces. The invention includes various embodiments of delay line and transmitter circuitries and methods that provide the series of pre-emphasis amplitude levels. For example, the invention advantageously includes both a synchronously clocked delay line and a master-slave calibrated delay line. Furthermore, both CML and LVDS transmitter circuitry implementations are provided. 
   The invention is advantageously applicable to both differential and single-ended signaling systems. Differential signaling involves the transmission of pairs of signals that propagate in parallel. Each is usually a logical complement of the other. That is, when one signal is at a high voltage (e.g., a logical 1), the other is at a low voltage (e.g., a logical 0), and vice versa. Pre-emphasis circuitry of the invention operates on the differential pair of signals substantially simultaneously. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The above and other advantages of the invention will be apparent upon consideration of the following detailed description, taken in conjunction with the accompanying drawings, in which like reference characters refer to like parts throughout, and in which: 
       FIG. 1  is a simplified block diagram illustrating circuitry involved in signal transmission that can be constructed in accordance with the invention; 
       FIG. 2  is a simplified schematic diagram of a component that can be included in the circuitry of  FIG. 1  and that can be constructed in accordance with the invention; 
       FIGS. 3–5  are plots of a signal transition at several points in the circuitry of  FIG. 1 ; 
       FIG. 6  is a plot of a pre-emphasized signal by known pre-emphasis circuitry; 
       FIG. 7  is a plot of a pre-emphasized signal by pre-emphasis circuitry constructed in accordance with the invention; 
       FIG. 8  is a simplified block diagram of an illustrative embodiment of pre-emphasis circuitry constructed in accordance with the invention; 
       FIG. 9  is a simplified block diagram of another illustrative embodiment of pre-emphasis circuitry constructed in accordance with the invention; 
       FIG. 10  is a simplified block diagram of an embodiment of delay line circuitry that can be used in the pre-emphasis circuitries of  FIGS. 8 and 9  in accordance with the invention; 
       FIG. 11  is a circuit diagram of an embodiment of a delay stage that can be used in the delay line circuitry of  FIG. 10  in accordance with the invention; 
       FIG. 12  is a block diagram of another embodiment of delay line circuitry that can be used in the pre-emphasis circuitries of  FIGS. 8 and 9  in accordance with the invention; 
       FIG. 13  is a circuit diagram of an embodiment of a delay stage that can be used in the delay line circuitry of  FIG. 12  in accordance with the invention; 
       FIG. 14  is a circuit diagram of an embodiment of a CML transmitter circuit that can be used in the pre-emphasis circuitry of  FIG. 9  in accordance with the invention; 
       FIG. 15  is a circuit diagram of an embodiment of a LVDS transmitter circuit that can be used in the pre-emphasis circuitry of  FIG. 9  in accordance with the invention; 
       FIG. 16  is a simplified block diagram illustrating circuitry that can be used in equalization/receiver circuitry in accordance with the invention; 
       FIG. 17  is a simplified block diagram of illustrative circuitry employing the invention; and 
       FIG. 18  is a simplified block diagram of an illustrative system employing the invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   An illustrative digital data transmission system is shown in  FIG. 1 . System  100  includes transmitter circuit  102 , communication link  103 , and receiver circuit  104 . In this embodiment, communication link  103  includes two conductors  103   a  and  103   b  (e.g., signal wires) extending from transmitter  102  to receiver  104 . Communication link  103  is a differential signaling link, which means that the value of a digital data bit is indicated by whether the voltage on conductor  103   a  is higher than the voltage on conductor  103   b  or vice versa. For example, a logical 1 data bit may be indicated by the voltage on conductor  103   a  being higher than the voltage on conductor  103   b , and a logical 0 data bit may be indicated by the voltage on conductor  103   b  being higher than the voltage on conductor  103   a . The signals on the two conductors are effectively complements of one another. Differential signaling has a number of protocols, such as, for example, LVDS and CML, to which system  100  can be designed to operate. The invention advantageously can support many of these protocols, whether industry standard, non-standard, or variations thereof. 
     FIG. 2  shows an output driver  200  that can be included in transmitter circuit  102 . Driver  200  converts a data signal (e.g., generated elsewhere in transmitter  102 ) to a form suitable for transmission on conductors  103   a,b . Driver  200  receives input signal VIN+, which can be considered a “true” version of the data signal, and input signal VIN−, which is a complement or inverted version of the data signal. (Alternatively, driver  200  may receive only one of these signals and may itself generate any necessary inverted version of the received signal.) Driver  200  responds to the VIN signals by producing signals VOUT+ and VOUT− on respective conductors  103   a,b . These VOUT signals represent the data content of the VIN signals and conform to the various parameters of the differential signaling protocol adhered to by communication link  103 . For example, these parameters may include maximum and minimum voltages, permissible common mode voltage, polarity of the voltage difference representing logical 1 and logical 0 data bits, etc. 
   Advantageously, output driver  200  can be designed in accordance with the invention to pre-emphasize signals VOUT+ and VOUT− corresponding to every transition of signals VIN+ and VIN−, respectively. This pre-emphasis is extra voltage applied to the VOUT+ and VOUT− signals for a certain amount of time immediately following every transition of the input signals. 
     FIGS. 3–5  show the effect of an ideal pre-emphasis in a logical 0 to logical 1 signal transition.  FIG. 3  shows an ideal voltage waveform (versus time) of an output signal as it is output from a transmitter circuit onto a communication link. Note the initial extra voltage of the signal above the logical 1 voltage level.  FIG. 4  shows an ideal voltage waveform of the communication link&#39;s response to receiving the output signal. Note the rapid rise time to the logical 1 voltage level.  FIG. 5  shows an ideal voltage waveform produced by a receiver circuit. Again, note the rapid rise time to the logical 1 voltage level. The degree to which an ideal pre-emphasis is achieved is determined in large part by the output generated by the pre-emphasis circuitry. 
     FIG. 6  shows the output of known pre-emphasis circuitry. Typically, the extra voltage of generated output  606  is applied during an entire baud period (T). A baud period can be simplistically thought of (for clarity with respect to the invention) as the minimum time period at which an input signal maintains the same value. Thus, for example, if input data consists of three sequentially transmitted logical bits “010,” the minimum period of time between the signal changing from 0 to 1 and from 1 to 0 is a baud period. Note that the time a signal maintains the same value can be longer than a baud period. For example, if the input data consists of bit values “011,” the logical 1 value remains the same for longer than a baud period. A disadvantage of known pre-emphasis circuitry is evident from the relatively large error between generated output  606  and the desired response  608 . A closer approximation to the desired response (i.e., an improved pre-emphasis) ultimately results in a stronger, more rapid transition at the receiver circuit. 
     FIG. 7  shows the generated output of one embodiment of pre-emphasis circuitry in accordance with the invention. Generated output  706  has a series of amplitude levels  706   a–d  extending over two baud periods ( 2 T) that better approximates the desired response  708 . Advantageously, the invention can achieve even closer approximations by increasing the number of amplitude levels and/or extending the output beyond two baud periods. 
   Pre-emphasis applied for the entire duration of a baud period as shown in  FIG. 6  clearly consumes more power than pre-emphasis of the invention in which an applied series of amplitude levels gradually decreases as shown in  FIG. 7 . Pre-emphasis applied for the entire duration of a baud period may also increase jitter, which is the abrupt, spurious variation in the magnitude of successive signal pulses. 
     FIG. 8  shows a high level pre-emphasis transmitter circuit in accordance with the invention. Pre-emphasis circuit  800  includes a delay line  810 , which has (n−1) delay elements  812 . Each delay element  812  has a unit delay equal to T/m, where T is a baud period and m is an integer. Pre-emphasis circuit  800  also includes n coefficient multiplication blocks  814 . Note that in known pre-emphasis circuits used in gigabit data transmission, m=1 and n=1. Pre-emphasis circuit  800  further includes analog adder  814  and transmitter driver  816 . Coefficient blocks  814 , delay line  810 , and adder  814  form a FIR (finite infinite response) filter. FIR filters are used to implement digitally many different types of output responses. In general, depending on the duration of each input and the total delay of the delay line, a FIR filter produces a weighted average of its n most recent inputs or a fractional weighted average of one input. Returning to circuit  800 , each of the inputs to adder  814  represents a “tap” of the filter, which when output, forms all or part of the amplitude levels shown in  FIG. 7 . The amplitude of each level is determined by the individual coefficients, and the duration of each amplified level is determined by the unit delay T/m of delay line  810 . Moreover, additional amplitude levels of finer granularity can be advantageously output by increasing the length of the FIR filter (i.e., by adding additional unit delays  812  and corresponding coefficient blocks  814 ). 
     FIG. 9  shows another embodiment of pre-emphasis circuitry in accordance with the invention. Pre-emphasis circuitry  900  includes delay line  910 , which has a number of delay blocks denoted Z-1 each having a delay of T/m. Circuitry  900  also includes output driver  920 , which may be, for example, a CML or LVDS protocol driver, and which may use differential (as shown) or single-ended signaling. The load may include, for example, a printed circuit board or backplane. Circuitry  900  further includes logic circuitry  922  and a number of current sources  924 . The number of current sources  924  is equal to the number of delay blocks in delay line  910 . Note that the four delay blocks and four current sources  924  shown in  FIG. 9  are merely illustrative, other numbers of delay blocks and current sources can be used depending on the desired number and duration of amplitude levels in the output. 
   An input data stream enters output driver  920  as well as delay line  910 . Output driver  920  initially outputs an amplified signal at a maximum voltage level. As the delay line outputs are fed to logic circuitry  922  after their respective delays are incurred, logic circuitry  922  generates activation or enablement signals for current sources  924 . As each current source  924  is in turn activated, more and more current is drawn from driver  920 , thus reducing the voltage level of the output signal in a series of steps. Each step has the approximate duration of a Z-1 delay block. The relative size of each current source  924  (i.e., the amount of current that each current source  924  can sink) can be selected as desired to give the output waveform desired amplitude levels. An output waveform identical or similar to  FIG. 7  can thus be created. Note that the peak-to-peak saturation voltages of the transistors used in driver  920  (see  FIGS. 14 and 15  for circuit implementations) are changed by the changing supply currents  11 – 14 . The output driver can thus be commonly used in a digital circuit regenerative type driver as opposed to a linear, unity gain driver. 
   Advantageously, the pre-emphasis circuits of  FIGS. 8 and 9  consume less power than known pre-emphasis circuits generating outputs having the waveform of  FIG. 6 . 
     FIG. 10  shows an embodiment of delay line circuitry that can be used to implement delay lines  810  and  910  in accordance with the invention. Delay line circuitry  1000  includes a series of alternating  1025   a  and  1025   b  delay stage D-latches. The 1025b D-latches receive an inverting clock input. The alternating use of inverting and non-inverting clock inputs results in a 1/2 clock period data delay between delay stages. The odd numbered stages (e.g., stage  1 , stage  3 , etc.) move data from input D to output Q on the rising edge of the clock signal, and the even numbered stages (e.g., stage  2 , stage  4 , etc.) move data on the falling edge of the clock signal. This arrangement can result in the generation of an output signal having two amplitude level steps per baud period over a given number of baud periods. 
     FIG. 11  shows a differential signaling CMOS circuit implementation for the D-latch of  FIG. 10  in accordance with the invention. D-latch  1125  is suitable for gigabit operation and includes resistors  1127  and  1129  coupled to power supply voltage VDD and NMOS transistors  1130 – 1136 . Transistor  1130  receives the DATA signal while transistor  1132  receives the CLOCK signal. VBIAS is a control or enablement signal that when high (e.g., a logical 1 voltage) allows D-latch  1125  to operate. To use D-latch  1125  as an odd numbered stage  1025   a , the Q output is coupled to the next delay stage. The Q output receives the value of the DATA signal on the rising edge of the CLOCK signal. To use D-latch  1125  as an even numbered stage  1025   b , the Q complement output is coupled to the next delay stage. The Q complement output receives the value of the DATA signal on the falling edge of the CLOCK signal. 
     FIG. 12  shows another embodiment of delay line circuitry that can be used to implement delay lines  810  and  910  in accordance with the invention. Delay line circuitry  1200  does not use a clock source synchronous to the data stream. The fractional delay (of the baud period) is obtained via a calibrated master-slave delay line arrangement. Delay line circuitry  1200  includes delay line slave  1240 , which has n stages of inverting delay blocks  1242 , and master loop  1250 . Master loop  1250  includes phase detector  1251 , charge pump &amp; loop filter  1253 , divide-by-n circuit  1255 , and ring oscillator  1260 . Ring oscillator  1260  includes n stages of inverting delay blocks  1262 . Delay blocks  1242  and  1262  are preferably identical in both number and construction. 
   A clock source is applied to phase detector  1251 , which generates an error signal that in phase-locked loop arrangements aligns the phases (and thus the frequencies) of the signals at the phase detector inputs. These two signals are the clock signal and the divided-by-n signal out of ring oscillator  1260 . This results in the clock signal period T equaling n periods of the ring oscillator and (n×1) delays of the individual delay stages of the oscillator ring, where 1 is the number of stages in ring oscillator  1260 . Because the same control signal adjusts the speed of both delay line slave  1240  and oscillator  1260 , the delay of the slave becomes calibrated and is thus ensured of being (n×1) times shorter than the period of the clock signal. Advantageously, this arrangement allows arbitrary fractions of the baud period to obtained. In particular, 1/2, 1/3, and 1/4 ratios can be obtained. The fraction determines the number of output signal amplitude levels that can be provided within a baud period. 
     FIG. 13  shows an embodiment of a CMOS circuit that can be used to implement delay block stage  1242  and  1262 . Circuit  1342 / 62  includes resistors  1327  and  1329  and NMOS transistors  1130 – 1132 . Transistor  1330  receives the DATA input, transistor  1131  receives complementary DATA input, and transistor  1132  receives the CONTROL SIGNAL input. When input DATA and CONTROL SIGNAL are both high and complement DATA signal is low, transistors  1330  and  1332  are ON (i.e., conducting), while transistor  1331  is OFF (i.e., non-conducting). Output Q is thus low, while output complement Q is high. 
     FIG. 14  shows a CML embodiment of a CMOS circuit that can be used to implement the transmitter portion (including output driver  920  and current sources  924 ) of pre-emphasis circuitry  900  in accordance with the invention. CML circuit  1400  includes n current blocks connected in parallel and may be referred to as having n-taps. Circuit  1400  includes resistors  1427  and  1429  (coupled to power supply voltage VDD), output nodes A and complement A, and current blocks  1420   a,b,n . The number of current blocks in CML circuit  1400  determines the number of amplitude levels in the output signal. 
   Main current block  1420   a  includes NMOS transistors  1421   a ,  1423   a , and  1424   a . Transistor  1424   a  is controlled by signal ACTa and sinks current Imain when it and one of transistors  1421   a  and  1423   a  are ON. Transistor  1421   a  receives input signal DATA, while transistor  1423   a  receives the complement of input signal DATA. This current block generally corresponds to output driver  920 . 
   Similarly, current block  1420   b  includes NMOS transistors  1421   b ,  1423   b , and  1424   b . Transistor  1424   b  is controlled by signal ACTb and sinks current I 1  when it and one of transistors  1421   b  and  1423   b  are ON. Transistor  1421   b  receives input signal (DATA)Z −1 , while transistor  1423   b  receives the complement of input signal (DATA)Z −1 . These data signals are the same as those received by current block  1420   a , but delayed by about one unit delay (such as, e.g., the delay of one Z-1 block of  FIG. 9 ). When this current block is active, a logical 1 output at either output A or output complement A is reduced in amplitude by a voltage equal to (current I 1 )×(resistor  1427  or  1429 ). 
   Current block  1420   n  is similar to the others and includes NMOS transistors  1421   n ,  1423   n , and  1424   n . Transistor  1424   n  is controlled by signal ACTn and sinks current In when it and one of transistors  1421   n  and  1423   n  are ON. Note that currents Imain, I 1 , and In are all preferably constant, but not necessarily equal. Transistor  1421   n  receives input signal (DATA)Z −n , while transistor  1423   n  receives the complement of input signal (DATA)Z −n . These data signals are the same as those received by current block  1420   a , but delayed by the total delay of a delay line (such as, e.g., as received from the last Z-1 delay block in delay line  910  of  FIG. 9 ). Current block  1420   n  further reduces the voltage at either output A or output complement A by an additional voltage equal to (current In)×(resistor  1427  or  1429 ). 
   Activation signals ACTb-n may be generated from control logic, such as, for example, logic circuitry  922 . Such control logic receives input from a delay line (such as, for example, delay lines  810  and  910 ), which receives data signals to be transmitted. Activation signals ACTb-n may additionally be derived from main current block activation signal ACTa in conjunction with inputs received by a delay line. 
   Note that for clarity in  FIG. 9 , individual delayed data signals (such as, e.g., (DATA)Z −1  and (DATA)Z −n  shown in  FIG. 14 ) are not shown connected from delay line  910  to output driver  920 . 
     FIG. 15  shows an LVDS embodiment of a CMOS circuit that can be used to implement the transmitter portion (including output driver  920  and current sources  924 ) of pre-emphasis circuitry  900  in accordance with the invention. LVDS circuit  1500  includes n current blocks connected in parallel and may be referred to as having n-taps. Current blocks  1520   a,b,n  are each coupled to power supply voltage VDD and are coupled to output nodes A and complement A, which have a load, shown as a resistor, coupled between them. As in circuit  1400 , the number of current blocks in CML circuit  1500  determines the number of amplitude levels in the output signal. Main current block  1520   a  includes NMOS transistors  1521   a  and  1523   a , PMOS transistors  1541   a  and  1543   a , and a pair of current sources/sinks Imain. Transistors  1541   a  and  1523   a  receive input signal DATA, while transistors  1543   a  and  1521   a  receive the complement of input signal DATA. Current blocks  1520   b  and  1520   n  are constructed similarly (reference numerals for some circuit elements are omitted for clarity), and receive signals DATA and complement DATA delayed by a correspondingly respective number of unit delays. That is, current block  1520   b  receives data signals delayed by one unit delay, while current block  1520   n  receives data signals delayed by n unit delays. No separate activate or enable signal is required to operate circuit  1500 . 
   In addition to pre-emphasis circuitry, the principles of the invention are also advantageously applicable to equalization circuitry. Equalization circuitry provides receiver circuitry with the capability of increasing the strength of a received signals, especially immediately adjacent any transitions in the received signals. The receiver circuitry can therefore more rapidly begin to respond to a change in the data being transmitted. This allows systems to be operated more rapidly, more reliably, at lower voltages, and/or with various combinations of these advantages employed to various different degrees. 
     FIG. 16  shows a generalized embodiment of equalization circuitry that can be included in a receiver circuit, such as, for example, receiver circuit  104  of  FIG. 1 . Equalization circuitry  1600  includes delay line  1610  and adder  1616 , which outputs an equalized signal. Delay line  1610  includes a number of delay units  1612  (the three delay units shown are merely illustrative; delay line  1610  may have other numbers of delay units  1612 ). Advantageously, both the CML based delay stages and the summing CML and LVDS arrangements previously described may be used to obtain the equalized output signal. 
   Although the circuitry of this invention has many other possible applications, one illustrative use is shown in  FIG. 17 . In  FIG. 17 , programmable logic device (“PLD”)  1700  is an integrated circuit, preferably an integrated circuit chip, that includes programmable logic circuitry  1710  and output driver circuitry  1720 . Output driver circuitry  1720  includes pre-emphasis circuitry in accordance with the invention. PLD  1700  may be field programmable, mask programmable, or programmable in any other way. It may be one-time-only programmable, or it may be reprogrammable. Programmable logic circuitry  1710  produces a data output signal on conductor  1730  that is applied to output driver circuitry  1720 . Circuitry  1720  converts this signal to differential output signals VOUT+ and VOUT−, with pre-emphasis, as described earlier in this specification. If only single-ended signaling is desired, only one or the other of VOUT+ or VOUT− is used as mentioned above. PLD  1700  is thus one illustrative embodiment of transmitter circuitry incorporating pre-emphasis circuitry in accordance with the invention. 
     FIG. 18  shows an illustrative larger context in which the invention may be employed. The invention can be used for driving one or more output signals from any one or more of elements  1700 ,  1840 ,  1850 ,  1860 , and  1870  out onto system bus or other interconnections  1880 . Although the invention is equally applicable in many other types of systems, illustrative system  1800  shown in  FIG. 18  may be generally described as a data processing system. 
   Data processing system  1800  may include one or more of the following components: PLD or other circuitry  1700  like that shown in  FIG. 17 , a processor  1840 , a memory  1850 , input/output (I/O) circuitry  1860 , and peripheral devices  1870 . These components are coupled together by a system bus or other interconnections  1880 , and are populated on a circuit board  1890  (e.g., a printed circuit board) that is contained in system  1800 . Communication among the various components shown in  FIG. 18 , and/or with external circuitry, may be of any known type to any desired extent. 
   System  1800  can be used in a wide variety of applications, such as computer networking, data networking, instrumentation, video processing, digital signal processing, or the like. Circuitry  1700  can be used to perform a variety of different logic functions. For example, circuitry  1700  can be configured as a processor or controller that works in cooperation with processor  1840 . Circuitry  1700  may also be used as an arbiter for arbitrating access to a shared resource in system  1800 . In yet another example, circuitry  1700  can be configured as an interface between processor  1840  and one of the other components of system  1800 . Still further, either processor  1840 , memory  1850 , or both may include pre-emphasis circuitry in accordance with the invention. Note that system  1800  is only exemplary and in no way should be construed to limit the true scope and spirit of the invention. 
   Thus it is seen that pre-emphasis circuitries and methods are provided. One skilled in the art will appreciate that the invention can be practiced by other than the described embodiments, which are presented for purposes of illustration and not of limitation, and the invention is limited only by the claims which follow.