Abstract:
The invention relates to systems and methods for calibrating and using resistance temperature detectors. In one embodiment, the system includes a calibration circuit comprising a resistance temperature detector in a bridge circuit with at least one potentiometer, and a programmable gain amplifier coupled to the bridge circuit. Embodiments of the invention further comprise methods for calibrating the bridge circuit and the programmable gain amplifier for use with the resistance temperature detector and methods for determining the self heating voltage of the bridge circuit.

Description:
BACKGROUND 
       [0001]    1. Field of Invention 
         [0002]    The present invention relates to microfluidic devices and temperature control of the microfluidic devices for performing biological reactions. In some embodiments, the present invention relates to systems and methods for calibrating and using a resistance temperature detector for use in a microfluidic device. 
         [0003]    2. Discussion of the Background 
         [0004]    The detection of nucleic acids is central to medicine, forensic science, industrial processing, crop and animal breeding, and many other fields. The ability to detect disease conditions (e.g., cancer), infectious organisms (e.g., HIV), genetic lineage, genetic markers, and the like, is ubiquitous technology for disease diagnosis and prognosis, marker assisted selection, identification of crime scene features, the ability to propagate industrial organisms and many other techniques. Determination of the integrity of a nucleic acid of interest can be relevant to the pathology of an infection or cancer. 
         [0005]    One of the most powerful and basic technologies to detect small quantities of nucleic acids is to replicate some or all of a nucleic acid sequence many times, and then analyze the amplification products. Polymerase chain reaction (PCR) is a well-known technique for amplifying DNA. With PCR, one can produce millions of copies of DNA starting from a single template DNA molecule. PCR includes phases of “denaturation,” “annealing,” and “extension.” These phases are part of a cycle which is repeated a number of times so that at the end of the process there are enough copies to be detected and analyzed. For general details concerning PCR, see Sambrook and Russell,  Molecular Cloning—A Laboratory Manual  (3rd Ed.), Vols. 1 -3, Cold Spring Harbor Laboratory, Cold Spring Harbor, N.Y. (2000);  Current Protocols in Molecular Biology,  F. M. Ausubel et al., eds., Current Protocols, a joint venture between Greene Publishing Associates, Inc. and John Wiley &amp; Sons, Inc., (supplemented through 2005) and  PCR Protocols A Guide to Methods and Applications,  M. A. Innis et al., eds., Academic Press Inc. San Diego, Calif. (1990). 
         [0006]    The PCR process phases of denaturing, annealing, and extension occur at different temperatures and cause target DNA molecule samples to replicate themselves. Temperature cycling (thermocyling) requirements vary with particular nucleic acid samples and assays. In the denaturing phase, a double stranded DNA (dsDNA) is thermally separated into single stranded DNA (ssDNA). During the annealing phase, primers are attached to the single stand DNA molecules. Single strand DNA molecules grow to double stranded DNA again in the extension phase through specific bindings between nucleotides in the PCR solution and the single strand DNA. Typical temperatures are 95° C. for denaturing, 55° C. for annealing, and 72° C. for extension. The temperature is held at each phase for a certain amount of time which may be a fraction of a second up to a few tens of seconds. The DNA is doubled at each cycle; it generally takes 20 to 40 cycles to produce enough DNA for the applications. To have good yield of target product, one has to accurately control the sample temperatures at the different phases to a specified degree. 
         [0007]    More recently, a number of high throughput approaches to performing PCR and other amplification reactions have been developed, for example, involving amplification reactions in microfluidic devices, as well as methods for detecting and analyzing amplified nucleic acids in or on the devices. Thermal cycling of the sample for amplification is usually accomplished in one of two methods. In the first method, the sample solution is loaded into the device and the temperature is cycled in time, much like a conventional PCR instrument. In the second method, the sample solution is pumped continuously through spatially varying temperature zones. See, for example, Lagally et al. ( Analytical Chemistry  73:565-570 (2001)), Kopp et al. ( Science  280:1046-1048 (1998)), Park et al. ( Analytical Chemistry  75:6029-6033 (2003)), Hahn et al. (WO 2005/075683), Enzelberger et al. (U.S. Pat. No. 6,960,437) and Knapp et al. (U.S. Patent Application Publication No. 2005/0042639). 
         [0008]    Many detection methods require a large number of copies (millions, for example) of the original DNA molecule, in order for the DNA to be characterized. Because the total number of cycles is fixed with respect to the number of desired copies, the only way to reduce the process time is to reduce the length of a cycle. Thus, the total process time may be significantly reduced by rapidly heating and cooling samples to process phase temperatures while accurately maintaining those temperatures for the process phase duration. 
         [0009]    Accordingly, what is desired is a system and method for rapidly and accurately changing process temperatures in PCR and thermal melt processes. 
       SUMMARY 
       [0010]    In one aspect, the present invention provides an improved tunable temperature measurement circuit. In some embodiments, the improved tunable temperature measurement circuit includes a source node maintained at a predetermined source voltage; a ground node maintained at a predetermined ground voltage; and a bridge circuit coupled to programmable gain instrumentation amplifier. In one embodiment, the bridge circuit comprises (1) a first resistance temperature detector connected between the source node and a first measurement node, (2) a first reference resistor connected between the first measurement node and the ground node, (3) a potentiometer (e.g. a programmable digital potentiometer) connected between the source node and a reference node, and (4) a scaling resistor connected between the reference node and the ground node. The programmable gain instrumentation amplifier may be connected so that a first input to the first programmable gain instrumentation amplifier is connected to the reference node, a second input to the first programmable gain instrumentation amplifier is connected to the first measurement node, and the output of the first programmable gain instrumentation amplifier is representative of the temperature sensed by the first resistance temperature detector. In some embodiments of the improved tunable temperature measurement circuit, one or more of the first reference resistor and the scaling resistor are also potentiometers. 
         [0011]    In some embodiments, the improved tunable temperature measurement circuit also includes a capacitor connected in parallel with the scaling resistor and/or a low-pass filter coupled to the output of the first programmable gain instrumentation amplifier. 
         [0012]    In some embodiments, the improved tunable temperature measurement circuit also includes a bypass circuit connected between the first measurement node and the ground node, wherein the bypass circuit comprises a bypass switch (e.g., a digital switch) in series with a bypass resistor. In some embodiments, the bypass circuit is configured to pulse width modulate a current passing through the first resistance temperature detector. 
         [0013]    In some embodiments, the improved tunable temperature measurement circuit also includes a power control circuit connected to the first measurement node, wherein the power control circuit comprises a bottom power switch connected between the measurement node and a bottom power node maintained at the predetermined source voltage, and a grounding switch connected in series with a bypass resistor between the measurement node and the ground node. In some embodiments, the tunable temperature measurement circuit may also include a shunt circuit connected between the reference resistor and the ground node, wherein the shunt circuit comprises a shunt switch in parallel with a shunt resistor. 
         [0014]    In some embodiments, the improved tunable temperature measurement circuit also includes: a selector switch disposed in between the first resistance temperature detector and the first measurement node; and one or more second resistance temperature detectors connected to the source node in parallel with the first resistance temperature detector. In these embodiments, the selector switch may be configured to connect one of the first resistance temperature detector and the one or more second resistance temperature detectors to the measurement node. 
         [0015]    In some embodiments, the improved tunable temperature measurement circuit also includes: a second resistance temperature detector connected between the source node and a second measurement node, a second reference resistor connected between the second measurement node and the ground; and a second programmable gain instrumentation amplifier. In these embodiments, a first input to the second programmable gain instrumentation amplifier is connected to the reference node, a second input to the second programmable gain instrumentation amplifier is connected to the second measurement node, and the output of the second programmable gain instrumentation amplifier is representative of the temperature sensed by the second resistance temperature detector. In some embodiments, the improved tunable temperature measurement circuit also includes a unity gain buffer, wherein the reference node is connected to the programmable gain instrumentation amplifiers via the unity gain buffer. 
         [0016]    In another aspect, the invention provides a method of calibrating the potentiometer in an improved tunable temperature measurement system including the improved tunable temperature measurement circuit. In some embodiments, the method of calibrating the potentiometer includes the steps of: (a) setting the resistance value of the potentiometer to a first resistance value; (b) setting the gain of the first programmable gain instrumentation amplifier to a first gain value; (c) measuring the voltage output from the first programmable gain instrumentation amplifier; (d) in the case that the measured voltage is above a predetermined target value (e.g., a value selected to maximize the signal to noise ratio in the output of the first programmable gain instrumentation amplifier), adjusting the resistance value of the potentiometer in a first direction; (e) in the case that the measured voltage is below the predetermined target value, adjusting the resistance value of the potentiometer in a direction opposite to the first direction; and (f) repeating steps (c) through (e) until the measured voltage from the first programmable gain instrumentation amplifier is equal to the predetermined target value. 
         [0017]    In some embodiments, the method of calibrating the potentiometer in an improved tunable temperature measurement system of also includes the steps of: (g) after performing step (f), storing the resistance value of the potentiometer in an electronic memory; (h) associating the stored resistance value with an identifier corresponding to the first resistance temperature detector; (i) repeating steps (a) through (h) for a plurality of resistance temperature detectors to create a plurality of associations between resistance temperature detectors and resistance values; (j) detecting the presence of one of the plurality of resistance temperature detectors; and (k) setting the resistance value of the potentiometer to the resistance value associated with the one of the plurality of resistance temperature detectors. In some embodiments, the step of detecting the presence of one of the plurality of resistance temperature detectors comprises reading a machine readable bar code or an RFID tag from a platform chip containing the one of the plurality of resistance temperature detectors. 
         [0018]    In another aspect, the invention provides a method of calibrating the self-heating properties of the improved tunable temperature measurement system. In some embodiments, the method of calibrating the self-heating properties includes: (a) setting the predetermined source voltage to a first source voltage value corresponding to a desired operational supply voltage; (b) setting the gain of the first programmable gain instrumentation amplifier to a first gain value corresponding to a desired operational gain value; (c) measuring the voltage output from the first programmable gain instrumentation amplifier; (d) determining a first ratio of the output from the first programmable gain instrumentation amplifier to the source node voltage multiplied by the gain of the first programmable gain instrumentation amplifier; (e) decreasing the predetermined source voltage to a new source voltage value; (f) measuring the voltage output from the first programmable gain instrumentation amplifier; (g) determining a new ratio of the output from the first programmable gain instrumentation amplifier to the measured source node voltage multiplied by the gain of the first programmable gain instrumentation amplifier; (h) determining an asymptote ratio by repeating steps (e) through (g) until the change of the new ratio determined at (g) between subsequent iterations is beneath a predetermined threshold; and (i) determining an operational self-heating voltage difference by multiplying the desired operational gain value by the difference between the first ratio and the asymptote ratio. 
         [0019]    In some embodiments of the method of calibrating the self-heating properties of the improved tunable temperature measurement system, steps (c) and (f) further comprise measuring the voltage at the source node; and steps (d) and (g) use the measured voltage at the source node as the source node voltage. 
         [0020]    In some embodiments of the method of calibrating the self-heating properties of the improved tunable temperature measurement system, step (e) further comprises increasing the gain of the first programmable gain instrumentation amplifier to a new gain value such that the product of the first source voltage value and the first gain value is equal to the product of the new source voltage value and the new gain value. 
         [0021]    In another aspect, the invention provides a method for performing thermal calibration of the improved tunable temperature measurement system comprising the steps of: (a) setting the predetermined source voltage to a desired operational supply voltage; (b) setting the gain of the first programmable gain instrumentation amplifier to a desired operational gain value; (c) bringing the resistance temperature detector to a known temperature (e.g. by utilizing an externally controlled heating device that has been independently calibrated such as a Peltier device or a resistive heater); (d) measuring a voltage output from the first programmable gain instrumentation amplifier; (e) storing the measured output voltage in an electronic memory in association with the known temperature; (f) repeating steps (c) through (e) to store a plurality of associations between known temperatures and corresponding measured output voltages; and (g) utilizing the stored associations to calibrate the circuit for thermal variations (e.g. by utilizing a look up table for the plurality of known temperatures or by calculating a suitable curve to interpolate output voltage between the known temperatures). 
         [0022]    In another aspect, the invention provides a system of controlling the temperature of a microfluidic device for performing biological reactions. In some embodiments, the system of controlling the temperature of a microfluidic device for performing biological reactions includes an improved tunable temperature measurement circuit comprising a source node maintained at a predetermined source voltage; a ground node maintained at a predetermined ground voltage; and a bridge circuit coupled to programmable gain instrumentation amplifier. The bridge circuit comprises (1) a first resistance temperature detector connected between the source node and a first measurement node, (2) a first reference resistor connected between the first measurement node and the ground node, (3) a potentiometer connected between the source node and a reference node, and (4) a scaling resistor connected between the reference node and the ground node. The programmable gain instrumentation amplifier may be connected so that a first input to the first programmable gain instrumentation amplifier is connected to the reference node, a second input to the first programmable gain instrumentation amplifier is connected to the first measurement node, and the output of the first programmable gain instrumentation amplifier is representative of the temperature sensed by the first resistance temperature detector. 
         [0023]    In some embodiments, the system of controlling the temperature of a microfluidic device for performing biological reactions includes a bridge adjustment controller configured to: (a) set the resistance value of the potentiometer to a first resistance value; (b) set the gain of the first programmable gain instrumentation amplifier to a first gain value; (c) measure the voltage output from the first programmable gain instrumentation amplifier; (d) in the case that the measured voltage is above a predetermined target value, adjust the resistance value of the potentiometer in a first direction; (e) in the case that the measured voltage is below the predetermined target value, adjust the resistance value of the potentiometer in a direction opposite to the first direction; and (f) repeat steps (c) through (e) until the measured voltage from the first programmable gain instrumentation amplifier is equal to the predetermined target value. 
         [0024]    In some embodiments, the system of controlling the temperature of a microfluidic device for performing biological reactions includes a self-heating calibration controller configured to: (a) set the predetermined source voltage to a first source voltage value corresponding to a desired operational supply voltage; (b) set the gain of the first programmable gain instrumentation amplifier to a first gain value corresponding to a desired operational gain value; (c) measure the voltage output from the first programmable gain instrumentation amplifier; (d) determine a first ratio of the output from the first programmable gain instrumentation amplifier to the source node voltage multiplied by the gain of the first programmable gain instrumentation amplifier; (e) decrease the predetermined source voltage to a new source voltage value; (f) measure the voltage output from the first programmable gain instrumentation amplifier; (g) determine a new ratio of the output from the first programmable gain instrumentation amplifier to the measured source node voltage multiplied by the gain of the first programmable gain instrumentation amplifier; (h) determine an asymptote ratio by repeating steps (e) through (g) until the change of the new ratio determined at (g) between subsequent iterations is beneath a predetermined threshold; and (i) determine an operational self-heating voltage difference by multiplying the desired operational gain value by the difference between the first ratio and the asymptote ratio. 
         [0025]    In some embodiments, the system of controlling the temperature of a microfluidic device for performing biological reactions includes a thermal calibration controller configured to: (a) set the predetermined source voltage to a desired operational supply voltage; (b) set the gain of the first programmable gain instrumentation amplifier to a desired operational gain value; (c) bring the resistance temperature detector to a known temperature; (d) measure a voltage output from the first programmable gain instrumentation amplifier; (e) store the measured output voltage in an electronic memory in association with the known temperature; (f) repeat steps (c) through (e) to store a plurality of associations between known temperatures and corresponding measured output voltages; and (g) utilize the stored associations to calibrate the circuit for thermal variations 
         [0026]    The above and other embodiments of the present invention are described below with reference to the accompanying drawings. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0027]    The accompanying drawings, which are incorporated herein and form part of the specification, illustrate various embodiments of the present invention. In the drawings, like reference numbers indicate identical or functionally similar elements. Additionally, the left-most digit(s) of a reference number identifies the drawing in which the reference number first appears. 
           [0028]      FIG. 1  is a block diagram of a system for performing PCR and thermal melt analysis. 
           [0029]      FIG. 2  is a diagram of a microfluidic chip. 
           [0030]      FIG. 3   a  is a circuit diagram illustrating an improved tunable temperature measurement circuit according to an embodiment of the invention. 
           [0031]      FIG. 3   b  is a circuit diagram illustrating a programmable gain instrumentation amplifier according to an embodiment of the invention. 
           [0032]      FIG. 3   c  is a circuit diagram illustrating a capacitor connected in parallel with the scaling resistor according to some embodiments of the invention. 
           [0033]      FIG. 3   d  is a circuit diagram illustrating a low pass filter applied to the output of the programmable gain instrumentation amplifier according to some embodiments of the invention. 
           [0034]      FIG. 4  is a circuit diagram illustrating a bypass circuit according to an embodiment of the invention. 
           [0035]      FIG. 5  is a circuit diagram illustrating a selector switch according to an embodiment of the invention. 
           [0036]      FIG. 6   a  is a circuit diagram illustrating an improved tunable temperature measurement circuit according to an embodiment of the invention. 
           [0037]      FIG. 6   b  is a circuit diagram illustrating a sensing portion of an improved tunable temperature measurement circuit with a plurality of measurement nodes according to an embodiment of the invention. 
           [0038]      FIG. 6   c  is a circuit diagram illustrating an amplification and measurement portion of an improved tunable temperature measurement circuit with a plurality of measurement nodes according to an embodiment of the invention. 
           [0039]      FIG. 6   d  is a circuit diagram illustrating a programmable gain instrumentation amplifier according to an embodiment of the invention. 
           [0040]      FIG. 6   e  is a circuit diagram illustrating a power supply circuit for a tunable temperature measurement circuit with a plurality of programmable gain instrumentation amplifiers according to an embodiment of the invention. 
           [0041]      FIG. 6   f  is a circuit diagram illustrating control circuitry for a bypass circuit that performs multiplex measurements according to an embodiment of the invention. 
           [0042]      FIG. 7   a  is a circuit diagram illustrating an improved tunable temperature measurement circuit with a unity gain buffer according to an embodiment of the invention. 
           [0043]      FIG. 7   b  is a circuit diagram illustrating a unity gain buffer according to an embodiment of the invention. 
           [0044]      FIG. 8  is a circuit diagram illustrating an improved tunable temperature measurement circuit according to an embodiment of the invention. 
           [0045]      FIG. 9  is a flow chart illustrating a bridge adjustment method in accordance with embodiments of the invention. 
           [0046]      FIG. 10   a  is a flow chart illustrating a self-heating calibration method in accordance with embodiments of the present invention. 
           [0047]      FIG. 10   b  is a flow chart illustrating a self-heating calibration method in accordance with embodiments of the present invention. 
           [0048]      FIG. 10   c  is a flow chart illustrating a self-heating calibration method in accordance with embodiments of the present invention. 
           [0049]      FIG. 11  is a graph illustrating the ratio V out /G(V cc ) for varying values of V cc  in a tunable temperature measurement circuit in accordance with an embodiment of the present invention. 
           [0050]      FIG. 12  is a flow chart depicting a thermal calibration method in accordance with embodiments of the present invention. 
       
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
       [0051]      FIG. 1  illustrates a functional block diagram of a system  100  for performing PCR and thermal melt analysis according to some embodiments of the invention. As illustrated in  FIG. 1 , system  100  may include a microfluidic device  102 . Microfluidic device  102  may include one or more microfluidic channels  104 . In the examples shown, device  102  includes two microfluidic channels, channel  104   a  and channel  104   b.  Although only two channels are shown in the exemplary embodiment, it is contemplated that device  102  may have fewer than two or more than two channels. For example, in some embodiments, device  102  includes eight channels  104 . 
         [0052]    Device  102  may include two DNA processing zones, a DNA amplification zone  131 , sometimes referred to herein as PCR zone  131 , and a DNA melting zone  132 . A DNA sample traveling through the PCR zone  131  may undergo PCR, and a DNA sample passing through melt zone  132  may undergo high resolution thermal melting. As illustrated in  FIG. 1 , PCR zone  131  includes a first portion of channels  104  and melt zone  132  includes a second portion of channels  104 , which is down stream from the first portion. 
         [0053]    In order to achieve PCR for a DNA sample flowing through the PCR zone  131 , the temperature of the sample must be cycled, as is well known in the art. Accordingly, in some embodiments, system  100  includes a temperature control system  120 . The temperature control system  120  may include a temperature sensor, a heater/cooler, and a temperature controller. In some embodiments, a temperature control system  120  is interfaced with a main controller  130  so that main controller  130  can control the temperature of the samples flowing through the PCR zone and the melting zone. 
         [0054]    Main controller  130  may be connected to a display device for displaying a graphical user interface. Main controller  130  may also be connected to user input devices which allow a user to input data and commands into main controller  130 . 
         [0055]    To monitor the PCR process and the melting process that occur in PCR zone  131  and melt zone  132 , respectively, system  100  may include an imaging system  118 . Imaging system  118  may include an excitation source, an image capturing device, a controller, and an image storage unit. Other aspects of a suitable system in accordance with some aspects of the invention are disclosed in U.S. patent application Ser. No. 11/770,869, incorporated herein by reference in its entirety. 
         [0056]      FIG. 2  illustrates another embodiment of a microfluidic device  202 . As shown in  FIG.2 , the microfluidic chip  202  may comprise a plurality of microfluidic channels  204   a - 204   h  that traverse heater/sensor regions  206 ,  208 . The plurality of microfluidic channels  204   a - 204   h  may be disposed in alignment with heaters/sensors in a first region  206  configured to perform PCR and may also be disposed in alignment with heaters/sensors in a second region  208  configured to perform high resolution thermal melting. In some embodiments, heaters/sensors in the heater/sensor regions  206 ,  208  comprise resistance temperature detectors (“RTD”s). The RTDs may be composed from a variety of materials, for example, the RTDs may be platinum, copper, and/or nickel RTDs. 
         [0057]    The temperature-dependent resistance Rh of an RTD may be related to its temperature by the linear approximation shown in Equation 1: 
         [0000]        Rh ( T )= Rh ( T   0 )·[1+α( T−T   0 )]  Equation 1 
         [0000]    In equation 1, Rh(T 0 ) is a known resistance of the RTD at a predetermined temperature T 0 , T is the current temperature of the RTD, and a is a linear temperature coefficient of resistance. 
         [0058]    As the temperature T of the RTD increases, its resistance Rh will also increase according to Equation 1. Therefore, an accurate measurement of Rh(T) can indicate the temperature T of the RTD as shown in Equation 2 below: 
         [0000]    
       
         
           
             
               
                 
                   T 
                   = 
                   
                     
                       T 
                       0 
                     
                     + 
                     
                       
                         
                           
                             Rh 
                              
                             
                               ( 
                               T 
                               ) 
                             
                           
                           
                             Rh 
                              
                             
                               ( 
                               
                                 T 
                                 0 
                               
                               ) 
                             
                           
                         
                         - 
                         1 
                       
                       α 
                     
                   
                 
               
               
                 
                   Equation 
                    
                   
                       
                   
                    
                   2 
                 
               
             
           
         
       
     
         [0059]    One method of measuring the resistance Rh of the RTD is with a simple voltage dividing circuit. In some embodiments of a voltage dividing circuit, the RTD is placed in series with a reference resistor having a resistance Ri. The reference resistor may comprise a single resistive element, or in some embodiments may comprise a plurality of discrete elements that, in combination, exhibit a relatively constant resistance value. The RTD in series with the reference resistor are connected between a known voltage difference (e.g., V cc −V G ), and a voltage measurement V i  is taken at a point between the RTD and the reference resistor. In this configuration, the resistance Rh of the RTD can be calculated with the following relations: 
         [0000]    
       
         
           
             
               
                 
                   
                     ( 
                     
                       
                         V 
                         i 
                       
                       - 
                       
                         V 
                         G 
                       
                     
                     ) 
                   
                   = 
                   
                     
                       ( 
                       
                         
                           V 
                           cc 
                         
                         - 
                         
                           V 
                           G 
                         
                       
                       ) 
                     
                     · 
                     
                       Ri 
                       
                         
                           Rh 
                            
                           
                             ( 
                             T 
                             ) 
                           
                         
                         + 
                         Ri 
                       
                     
                   
                 
               
               
                 
                   Equation 
                    
                   
                       
                   
                    
                   3 
                 
               
             
             
               
                 
                   
                     Rh 
                      
                     
                       ( 
                       T 
                       ) 
                     
                   
                   = 
                   
                     Ri 
                     · 
                     
                       [ 
                       
                         
                           
                             ( 
                             
                               
                                 V 
                                 cc 
                               
                               - 
                               
                                 V 
                                 G 
                               
                             
                             ) 
                           
                           
                             ( 
                             
                               
                                 V 
                                 i 
                               
                               - 
                               
                                 V 
                                 G 
                               
                             
                             ) 
                           
                         
                         - 
                         1 
                       
                       ] 
                     
                   
                 
               
               
                 
                   Equation 
                    
                   
                       
                   
                    
                   4 
                 
               
             
           
         
       
     
         [0000]    By substituting the expression for Rh from equation 4 into equation 2, one can determine the temperature of the RTD by measuring the voltage V i : 
         [0000]    
       
         
           
             
               
                 
                   T 
                   = 
                   
                     
                       T 
                       0 
                     
                     + 
                     
                       
                         
                           
                             Ri 
                             
                               Rh 
                                
                               
                                 ( 
                                 
                                   T 
                                   0 
                                 
                                 ) 
                               
                             
                           
                           · 
                           
                             [ 
                             
                               
                                 
                                   ( 
                                   
                                     
                                       V 
                                       cc 
                                     
                                     - 
                                     
                                       V 
                                       G 
                                     
                                   
                                   ) 
                                 
                                 
                                   ( 
                                   
                                     
                                       V 
                                       i 
                                     
                                     - 
                                     
                                       V 
                                       G 
                                     
                                   
                                   ) 
                                 
                               
                               - 
                               1 
                             
                             ] 
                           
                         
                         - 
                         1 
                       
                       α 
                     
                   
                 
               
               
                 
                   Equation 
                    
                   
                       
                   
                    
                   5 
                 
               
             
           
         
       
     
         [0000]    Accordingly, one can accurately determine the temperature T of the RTD when Rh(T 0 ), α, V cc , and V G  are known. 
         [0060]    It is desirable to maximize the sensitivity of the measurement to changes in temperature (i.e., maximize ΔV i /ΔT). The sensitivity is maximized when the resistance of the RTD is equal to the resistance of the reference resistor. Accordingly, the nominal value for the resistance Ri of the reference resistor preferably should be selected to be approximately equal to the resistance Rh of the RTD over the expected range of temperatures in which the measurement circuit will be used. 
         [0061]    To better utilize the voltage range of the measurement V i  and to improve the signal to noise ratio of the temperature detection, in accordance with one embodiment the RTD may be placed within a bridge circuit.  FIG. 3   a  illustrates a circuit diagram for a tunable temperature measurement circuit  300  in accordance with one embodiment of the present invention. As shown in  FIG. 3   a,  the tunable temperature measurement circuit  300  may comprise a source node  303  maintained at a predetermined source voltage (V cc ), a ground node  304  maintained at a predetermined ground voltage (V G ), and a bridge circuit  301  coupled to a differential amplifier  302 , which in some embodiments may be a programmable gain instrumentation amplifier. Differential amplifier  302  is referred to herein as a programmable gain instrumentation amplifier, but the differential amplifier in some embodiments need not be limited to a programmable gain instrumentation amplifier. In one embodiment, the bridge circuit  301  may comprise a first leg (“measurement leg”)  301   a  including an RTD  305  having a temperature-dependent resistance Rh(T) connected between the source node  303  and a first measurement node  306  and a reference resistor  307  having a resistance Ri connected between the first measurement node  306  and the ground node  304 . The bridge circuit may also comprise a second leg (“reference leg”)  301   b  including a potentiometer  308  having an adjustable resistance Rj connected between the source node  303  and a reference node  309 , and a scaling resistor  310  having a resistance Rk connected between the reference node  309  and the ground node  304 . In some embodiments, potentiometer  308  is a programmable digital potentiometer. 
         [0062]    In one exemplary embodiment, the relative resistances in the bridge circuit  301  are selected to be proportional so that: 
         [0000]        Rj=C·Rh ( T   0 )   Equation 6 
         [0000]        Rk=C·Ri    Equation 7 
         [0000]    In the above relations, C is a scaling factor that ensures the current through the second leg is small, which in turn ensures that the potentiometer  308  and the scaling resistor  310  do not heat significantly. In one embodiment, this scaling factor may be in a range between 1 and 1000, and is preferably 100. 
         [0063]    In temperature measurement circuit  300 , the voltage difference between measurement node  306  (V i ) and reference node  309  (V k ) are compared via instrumentation amplifier  302  to output a precise signal corresponding to the temperature sensed by RTD  305 . 
         [0064]      FIG. 3   b  illustrates one embodiment of a programmable gain instrumentation amplifier  302 . As shown in  FIG. 3   b,  the relationship between the voltage out V and the difference between the input voltages V i  and V k  can be controlled by adjusting potentiometer R 1 . In some embodiments, potentiometer R 1  is a programmable digital potentiometer. 
         [0065]    In the embodiment shown in  FIG. 3   a,  the voltage V i  of the measurement node  306  is coupled to the non-inverting input of the programmable gain instrumentation amplifier  302 , while the voltage V k  from the reference node  309  is coupled to the inverting input of the programmable gain instrumentation amplifier  302 . In alternate embodiments, the voltage V i  of the measurement node  306  may be coupled to the inverting input of the programmable gain instrumentation amplifier  302 , while the voltage V k  from the reference node  309  is coupled to the non-inverting input of the programmable gain instrumentation amplifier  302 . 
         [0066]    Referring to  FIGS. 3   c  and  3   d,  certain features contained in some embodiments of a temperature measurement circuit  300  are illustrated. As shown in  FIG. 3   c,  the reference leg  301   b  of the bridge circuit  301  may also comprise a capacitor  311  connected in parallel with the scaling resistor  310 . This capacitor  311  tends to make the reference side of the bridge circuit  301  more stable and less prone to high frequency noise. As shown in  FIG. 3   d,  a low pass filter may be coupled to the output of the instrumentation amplifier  302 . The filter may comprise a resistor  312  and a capacitor  313 . The resistance of the resistor  312  and the capacitance of the capacitor  313  are selected to filter out high frequency noise such as, for example, variations at a rate exceeding approximately 10 kHz. In some embodiments the resistance of the resistor  312  may be in a range between 200Ω and 10,000Ω, and is preferably 820Ω. In other embodiments, the capacitance of the capacitor  313  may be in a range between 100 pF and 40,000 pF, and is preferably 500 pF. 
         [0067]      FIG. 4  illustrates a tunable temperature measurement circuit  300  in accordance with other embodiments. As shown in  FIG. 4 , the temperature measurement circuit  300  may further comprise a bypass circuit  400  for providing greater control over the amount of current passing through the RTD  305 . The bypass circuit  400  may include a switch  410 , such as a digital switch, and a low resistance resistor  411  having resistance Rd. The low resistance resistor  411  is switched into the circuit to maximize the current flowing through the RTD  305 . In some embodiments the resistance of the low resistance resistor  411  may be in a range between 0Ω and 1,000Ω, and is preferably 0Ω. In other embodiments, the resistance of the low resistance resistor  411  is selected to be substantially smaller than the resistance Ri of the reference resistor  307 . In some embodiments, switch  410  is used in conjunction with pulse width modulation to enable greater control over the current through RTD  305 . 
         [0068]    Referring to  FIG. 5 , another embodiment of the tunable temperature measurement circuit  300  is illustrated. As shown in  FIG. 5 , the temperature measurement circuit may be configured to accommodate a plurality of RTDs (e.g. RTDs  505   a  and  505   b ) using switching circuitry. The plurality of RTDs may be configured to measure the temperatures of different environments or may be selected to possess different temperature-dependent resistances appropriate for measuring different ranges of temperatures. In some embodiments, the switching circuitry may comprise a selector switch  512 . As shown in  FIG. 5 , the selector switch  512  can be configured to connect one of the RTDs  505   a,    505   b  to the measurement node  306 . The potentiometer  308  and programmable gain instrumentation amplifier  302  may be adjusted to accommodate each RTD as it is switched into the circuit. 
         [0069]      FIG. 6   a  illustrates yet another embodiment of temperature detector circuit  300 . As shown in  FIG. 6   a,  the bridge circuit may be configured to simultaneously measure the resistance of a plurality of RTDs by providing one or more additional measurement legs  601 . In some embodiments, the additional measurement leg  601  may comprise an additional RTD  605  and an additional reference resistor  607 . Additionally, the embodiment shown in  FIG. 6   a  includes an additional programmable gain instrumentation amplifier  602  coupled to a measurement node  606  of the additional measurement leg  601 . In this embodiment, reference node  309  may be coupled to the plurality of the programmable gain instrumentation amplifiers  302 ,  602  and thus a single reference node  309  may provide a reference voltage for the plurality of RTDs  305 ,  605 . 
         [0070]    Referring to  FIG. 6   b,  a relationship between a microfluidic chip  102 ,  202  and the measurement legs  601  of the temperature detecting circuit  300  according to one embodiment is illustrated. As shown in  FIG. 6   b,  a plurality of measurement legs  601  (e.g.  601   a  through  601   h ) may be incorporated into a microfluidic chip  102 ,  202  while the remaining portion of the temperature detecting circuit is incorporated into a temperature control system, e.g. temperature control system  120 . This configuration can enable a single temperature measurement circuit to be used with a plurality of distinct microfluidic chips and can also simplify the fabrication of the microfluidic chips by reducing the circuitry thereon. 
         [0071]      FIG. 6   c  illustrates an exemplary embodiment of the programmable gain instrumentation amplifiers coupled to the measurement legs  601   a  though  601   h  shown in  FIG. 6   b.  As shown in  FIG. 6   c,  the measurement node (AI- 0  through AI- 7 ) for each measurement leg  601   a - 601   h  is connected to a separate programmable gain instrumentation amplifier  602   a  through  602   h.  Furthermore, as shown in  FIG. 6   c,  the voltage V k  of the reference node  309  is shared among all of the programmable gain instrumentation amplifiers in accordance with one embodiment. As described with reference to  FIG. 6   b,  in some embodiments the programmable gain instrumentation amplifiers  602   a  through  602   h  and associated control circuitry may be incorporated into a temperature control system  120  rather than integrated with the microfluidic chip  102 ,  202 . As illustrated in  FIG. 6   d,  in some embodiments, the programmable gain instrumentation amplifiers may be implemented using, for example, Single Resistor Gain Programmable, Precision Instrumentation Amplifiers, Linear Technology part no. LT1167. 
         [0072]    Referring to  FIGS. 6   e  and  6   f,  schematics of circuitry for performing a multiplex measurement technique according to some embodiments of the invention are illustrated. In one embodiment, the multiplex measurement technique involves switching a common power supply circuit  650  and a plurality of switching circuits  651  (see  FIG. 6   f ) to operate each RTD measurement leg (e.g.  601   a  through  601   h ) independently. 
         [0073]      FIG. 6   e  illustrates the common power supply circuit  650 . Power supply circuit  650  may comprise electric switches  652 , which can be Metal Oxide Semiconductor Field Effect Transistors (MOSFET) switches, that are driven by a digital line “Top Power.” The switches  652  may be used to connect or disconnect a top power source  653  (e.g. +30V) to a common lead  654 . 
         [0074]      FIG. 6   f  illustrates one embodiment of the switching circuit  651  for one of the eight RTD measurement legs (e.g., the measurement leg  601   a  corresponding to RTD  605   a ). The switching circuits  651  may include electric switches  655 , which can be MOSFET switches, that are driven by a digital line “Bottom Power,” for connecting the corresponding RTD (e.g. RTD  605   a ) with a bottom power source  656  (e.g. +30V). 
         [0075]    The switching circuits  651  may also include electric switch  657 , which can be a MOSFET switch, that is driven by a digital line “Heater,” for connecting the corresponding RTD (e.g.  605   a ) with a reference voltage  658  (e.g. 0 V) via a low resistance resistor  659  having a relatively a low resistance (e.g. 0Ω). In some embodiments, the resistance of the low resistance resistor  659  may be in a range between 0Ω and 1,000Ω, and is preferably 0Ω. The low resistance resistor  659  is switched into the circuit to maximize the current flowing through the RTD  605   a  and cause the RTD  605   a  to heat rapidly. 
         [0076]    The switching circuits  651  may also include electric switch  660 , which can be a MOSFET switch, that is driven by a digital line “Measure,” for connecting the corresponding RTD (e.g.  605   a ) with the reference voltage  658  via a reference resistor  607   a  having a relatively high resistance (e.g. 1,000Ω). In some embodiments, the resistance of the low resistance resistor  659  is selected to be substantially smaller than the resistance of the reference resistor  607   a.    
         [0077]    Additionally, as illustrated in  FIG. 6   f,  in some embodiments, the switching circuits  651  may include a shunt resistor  661  connected in parallel with the switch  660  and having a resistance value substantially higher than the reference resistor  607   a  (e.g. 1,000,000Ω). The shunt resistor  661  acts as a shunt around switch  660  when switch  660  is OFF. With the switch  660  ON, the resistance measurements can be taken as normal. When the switch  660  is OFF, however, then resistance measurements can still be taken due to the small current that still flows through the shunt resistor  661 . 
         [0078]    In one embodiment, each of the remaining RTD measurement legs (e.g.  601   b - 601   h ) also includes a switching circuit  651 . With the circuit illustrated in  FIGS. 6   e  and  6   f,  the common lead  654  can be disconnected from the top power source  653 , each RTD can be selectively connected to the bottom power source  656  and reference voltage  658 , and each RTD can be selectively removed from the resistive network. This embodiment thus allows for isolated, power-on and power-off measurements. 
         [0079]    In some embodiments, the circuit illustrated in  FIGS. 6   e  and  6   f  may be used to measure the series resistance across any two of the RTDs (e.g.,  605   a - 605   h ). For example, the Top Power signal may be used to set the switches  652  to the OFF state and disconnect the top power source  653  from the common lead  654 . Next, the Bottom Power signal corresponding to a first switching circuit  651  may be used to connect the bottom power source  656  to the corresponding first RTD (e.g.,  605   a ). Then, the Measure signal of a second switching circuit  651  may be used to connect the reference voltage  658  to the corresponding second RTD (e.g.,  605   b ) via the corresponding reference resistor (e.g.  607   b ). These settings will cause current to flow from the bottom power source  656 , through the first RTD (e.g.,  605   a ), the second RTD (e.g.,  605   b ), and the reference resistor (e.g.  607   b ) in series to the reference voltage  658 , and the voltage at the measurement node (e.g. AI- 1 ) of the second switching circuit  651  will correspond to the series resistance of the first and second RTDs (e.g.  605   a  and  605   b ). By controlling the switches of each of the switching circuits  651 , many other combinations of resistances can be measured. Aspects of additional circuitry for performing multiplex measurement techniques are disclosed in commonly assigned U.S. patent application Ser. No. 12/165,043, incorporated herein by reference in its entirety. 
         [0080]    In some embodiments wherein multiple RTDs share a common reference node, the multiplex measurement technique described above can create undesirable cross-talk. In some embodiments, the RTD leads fluctuate between three different voltage levels: V cc , V G  ground, and V measure . These three states occur when the RTD is used as the power supply side for multiplex measurement, when it is heated, and when it is used as a sensor, respectively. Using the multiplex circuit technique, only one channel will be at V measure  at a given moment and the other channels will be at a voltage close to V cc . However, whenever the voltage of a channel goes to V cc  or ground, this can force the instrumentation amplifier associated with that particular sensor into saturation (also called the overload condition). Using the shared reference node configuration, this overload of one instrumentation amplifier may affect the reference voltage V k , causing one sensor&#39;s overload to modify another sensor&#39;s reference voltage. 
         [0081]      FIG. 7   a  illustrates an embodiment of the temperature detecting circuit  300  that addresses this cross-talk issue. As shown in  FIG. 7   a,  a unity gain buffer  720  is placed in between the reference node  309  and the programmable gain instrumentation amplifiers  302 ,  602 . This buffer reduces the effect of cross-talk between the sensors by preventing the overload condition of one instrumentation amplifier from affecting the reference voltage V k . 
         [0082]      FIG. 7   b  illustrates a unit gain buffer  720  in accordance with one embodiment. As shown in  FIG. 7   b,  a unity gain buffer receives input voltage V k  from the reference node and outputs voltage V k ′ to the programmable gain instrumentation amplifiers. The low output impedance of the unit gain buffer  720  provides the current required by any saturated instrumentation amplifiers and allows the unsaturated amplifiers to work as designed without affecting the reference voltage. 
         [0083]    Referring to  FIG. 8 , another embodiment of the temperature detecting circuit  300  is illustrated. As shown in  FIG. 8 , in this embodiment, the bridge circuit  301  is fully adjustable with programmable resistances. In addition to potentiometer  308 , the reference resistor  807  and scaling resistor  810  may also comprise potentiometers. In some embodiments, the potentiometers may be programmable digital potentiometers. An advantage of this configuration is its ability to accept widely varying heater/sensor Rh resistances while optimizing the signal to noise ratio. With this configuration it is possible for a single temperature controller  120  to accept different platform chips  102 ,  202  that perform different biological assays. Some assays may require a different PCR protocol with additional thermocycling (e.g. small amplicon or probe melting). The added thermocycling may be achieved with a longer microchannel and correspondingly longer thin-film heater/sensor Rh. The longer heater/sensor Rh would then likely have a significantly larger resistance. By adding programmable reference resistor  807 , improved sensitivity can be maintained, such as, for example, when Rh=Ri. In some embodiments, programmable resistor  810  is included to maintain a fixed bridge ratio C even while the resistance Ri of the reference resistor  807  varies. 
         [0084]    In one exemplary embodiment, if the RTDs in platform chip “A” have resistances of 100Ω and the RTDs in platform chip “B” have resistances of 250Ω, then the programmable reference resistor  807  could be digitally adjusted to 100Ω or 250Ω as required. Of course, there is no limit to the number of different platform chips that could be used because the reference resistor  807  and the scaling resistor  810  are programmed as required. Furthermore, a single platform chip could have two or more very different RTD resistances. This may improve the functionality of the device by enabling two or more very different kinds of assays on the same chip (i.e. a hybrid chip that simultaneously runs two very different PCR protocols). In this case, the resistors  807  and  810  would simply be programmed different for each RTD. 
         [0085]    Finally, this fully adjustable bridge configuration could be used along with the shared bridge configuration discussed above to create a highly flexible, but still efficient, measurement system. 
         [0086]    Bridge Adjustment 
         [0087]    In some embodiments, the temperature-dependent resistance Rh(T) of the RTD may vary due to manufacturing variations, contact irregularities, corrosion, differences in design, etc. In a bridge adjustment process the potentiometer  308  in the tunable temperature measurement circuit  300  is tuned to account for variable RTD characteristics. 
         [0088]      FIG. 9  illustrates a flow chart describing a bridge adjustment process  900  for adjusting the potentiometer  308  in the temperature measurement circuit  300  in accordance with another aspect of the present invention. In some embodiments, the bridge adjustment process  900  may be performed by a bridge adjustment controller. Process  900  may begin at  920 , where the bridge adjustment controller may set an initial resistance value for the potentiometer  308 . At step  930 , the bridge adjustment controller may set an initial gain value G for the programmable gain instrumentation amplifier  302 . 
         [0089]    After these initial values are set at steps  920  and  930 , the bridge adjustment controller measures the output voltage V out  in step  940 . As discussed above, the output voltage will be indicative of the gain value G multiplied by the difference between the voltage V i  of the measurement node  306  and the voltage V k  of the reference node  309 . For example, in embodiments where the measurement node  306  is coupled to the non-inverting input of the programmable gain instrumentation amplifier  302 , and the reference node  309  is coupled to the inverting input of the programmable gain instrumentation amplifier  302 , the output voltage V out  may follow the relation shown in Equation 8: 
         [0000]        V   out   =G ·( V−V   k )   Equation 8 
         [0090]    At step  950 , the bridge adjustment controller compares the output voltage V out  against a predetermined target voltage. 
         [0091]    In the case that the output voltage V out  is above the target voltage, at step  960  the bridge adjustment controller adjusts the resistance of the potentiometer in a first direction. For example, in embodiments where the measurement node  306  is coupled to the non-inverting input of the programmable gain instrumentation amplifier  302 , and the reference node  309  is coupled to the inverting input of the programmable gain instrumentation amplifier  302 , a voltage V out  above the target voltage may indicate that the voltage V k  at the reference node  309  is too low and the bridge adjustment controller decreases the resistance of the potentiometer  308  in order to increase the voltage V k  at the reference node  309 . 
         [0092]    In the case that the output voltage V out  is below the target voltage, at step  970  the bridge adjustment controller adjusts the resistance of the potentiometer in a second direction. For example, in embodiments where the measurement node  306  is coupled to the non-inverting input of the programmable gain instrumentation amplifier  302 , and the reference node  309  is coupled to the inverting input of the programmable gain instrumentation amplifier  302 , a voltage V out  below the target voltage may indicate that the voltage V k  at the reference node  309  is too high and the bridge adjustment controller increases the resistance of the potentiometer  308  in order to decrease the voltage V k  at the reference node  309 . 
         [0093]    In the case that the output voltage V out  is about equal to the target voltage (that is, the voltage difference between the reference node  309  and the measurement node  306  is within a predetermined margin), then the bridge adjustment controller may terminate the bridge adjustment process  900 . 
         [0094]    In some preferred embodiments, the predetermined target voltage is selected to utilize more of the range of the programmable gain instrumentation amplifier  302  and improve the signal to noise ratio. For example, a target output voltage of zero (0) volts may be selected to maximize the signal with respect to the common mode voltage (i.e. to maximize ΔV out /V i ). Common target output voltage ranges include 0 to +10 V, −5 to +5 V, and −10 to +10 V. In some preferred embodiments, the target output voltage is at the lowest voltage at the lowest temperature and the highest voltage at the highest temperature (or vice versa). 
         [0095]    In some embodiments, the RTD  305  may be integrated with a platform chip (e.g. the microfluidic device  202 ), while the potentiometer  308  may be integrated into a temperature control system (e.g. the temperature control system  120 ). In this embodiment, many distinct microfluidic devices or other temperature-controlled devices may be used with the same control system  120 . Each microfluidic device  202  may be marked with a machine readable identification, e.g., a machine readable bar code or a radio-frequency identification (“RFID”) tag. The temperature control system may read the machine readable identification and store the calibrated potentiometer setting for each device in association with the identification for that device. The temperature control system may also be configured to detect the machine readable identification of a device and program the potentiometer for that device based upon the previously stored settings. 
         [0096]    Self-Heating Calibration 
         [0097]    According to another aspect of the present invention, a self-heating calibration process is used to account for undesirable self-heating effects of the RTD. 
         [0098]    Using the RTD  305  as a temperature sensor requires passing an electrical current I through it. According to Joule&#39;s first law, an electrical current I passing through a resistor having resistance Rh(T) will dissipate an amount of power P as heat: 
         [0000]        P=I   2   ·Rh ( T )   Equation 9 
         [0000]    The heating from P will cause the RTD to rise in temperature by an amount ΔT above the actual environmental temperature. This increase in temperature above the environmental temperature is known as self-heating and can cause undesirable errors in measurement values. The specific magnitude of the temperature increase will depend upon the rate at which heat is being produced and the thermal resistance θ between RTD  305  and the environment: 
         [0000]      Δ T=θ·P    Equation 10 
         [0000]    Thus, the RTD  305  itself will be at a temperature ΔT higher than the environment. In general, ΔT is independent of the ambient temperature. For example, if ΔT is 5° C. and the environment is 20° C., then the temperature of RTD  305  will be 25° C. In an otherwise comparable environment (that is, in an environment where θ has not changed significantly) at 100° C., the temperature of the RTD  305  will be 105° C. 
         [0099]    The undesirable effects of self-heating can be minimized by limiting the current through the RTD  305 . This can be accomplished by increasing the resistance Ri of the reference resistor  307  or by reducing the supply voltage V cc . However, during operation, both of these changes could have undesirable consequences. As noted above, it is desirable in certain embodiments to match the resistance Ri of the reference resistor  307  to the resistance Rh of the RTD  305  in order to increase the sensitivity of the system. While increasing the resistance Ri of the reference resistor  307  would reduce the size of the self-heating effect ΔT, it would also reduce the sensitivity of the measurement leg of the bridge circuit  301 . Furthermore, a high supply voltage V cc  is often desired for a good common mode signal as well as for the ability to rapidly heat the RTD. 
         [0100]    Accordingly, it may be preferable to calibrate temperature measurement systems by calculating the self-heating voltage change ΔV sh  that will occur on the output voltage V out  under normal conditions (e.g., the desired resistance Ri of the reference resistor  307 , the desired gain setting G for the programmable gain instrumentation amplifier  302 , and the desired supply voltage V cc ) and simply remove this known error from measurements. 
         [0101]    According to some embodiments, the self-heating calibration process may comprise identifying voltage settings at which self-heating is minimized; measuring the resistance Rh of the RTD when self-heating is minimized; and comparing that value with the resistance Rh of the RTD under voltage settings in which self-heating is present. As described above and with reference to Equation 1, in the absence of self-heating an RTD may behave as an ohmic device (i.e., exhibit a linear I-V curve) if the ambient temperature T is held constant. Accordingly, in some embodiments the voltage settings at which self-heating is minimized can be indirectly determined by identifying the voltage settings at which the resistance Rh of the RTD remains relatively constant with respect to changes in voltage, that is, the voltage settings as which ΔRh/ΔV cc ≈0. As explained below, in some embodiments these voltage settings can also be identified as settings at which ΔV out /(G·ΔV cc ) remains constant. 
         [0102]    The voltage difference between the measurement node  306  and the reference node  309  (V i −V k ) is proportional to the voltage difference between the source voltage and the ground voltage (V cc −V G ), as illustrated in Equation 11: 
         [0000]    
       
         
           
             
               
                 
                   
                     ( 
                     
                       
                         V 
                         i 
                       
                       - 
                       
                         V 
                         k 
                       
                     
                     ) 
                   
                   = 
                   
                     
                       ( 
                       
                         
                           V 
                           cc 
                         
                         - 
                         
                           V 
                           G 
                         
                       
                       ) 
                     
                     · 
                     
                       ( 
                       
                         
                           Ri 
                           
                             Rh 
                             + 
                             Ri 
                           
                         
                         - 
                         
                           Rk 
                           
                             Rj 
                             + 
                             Rk 
                           
                         
                       
                       ) 
                     
                   
                 
               
               
                 
                   Equation 
                    
                   
                       
                   
                    
                   11 
                 
               
             
           
         
       
     
         [0000]    Setting the gain of the programmable gain instrumentation amplifier  302  to a value G causes the output voltage V out  to be proportional to the voltage difference between the measurement node  306  and the reference node  309  (V i −V k ), as illustrated in Equation 12: 
         [0000]        V   out   =G ·( V   i   −V   k )   Equation 12 
         [0000]    Substituting the expression in Equation 11 for (V i −V k ) into Equation 12 produces a relation between V out  and (V out −V G ), as shown below in Equations 13 and 14: 
         [0000]    
       
         
           
             
               
                 
                   
                     V 
                     out 
                   
                   = 
                   
                     G 
                     · 
                     
                       ( 
                       
                         
                           V 
                           cc 
                         
                         - 
                         
                           V 
                           G 
                         
                       
                       ) 
                     
                     · 
                     
                       ( 
                       
                         
                           Ri 
                           
                             Rh 
                             + 
                             Ri 
                           
                         
                         - 
                         
                           Rk 
                           
                             Rj 
                             + 
                             Rk 
                           
                         
                       
                       ) 
                     
                   
                 
               
               
                 
                   Equation 
                    
                   
                       
                   
                    
                   13 
                 
               
             
             
               
                 
                   
                     ρ 
                     ≡ 
                     
                       
                         V 
                         out 
                       
                       
                         G 
                         · 
                         
                           ( 
                           
                             
                               V 
                               cc 
                             
                             - 
                             
                               V 
                               G 
                             
                           
                           ) 
                         
                       
                     
                   
                   = 
                   
                     ( 
                     
                       
                         Ri 
                         
                           Rh 
                           + 
                           Ri 
                         
                       
                       - 
                       
                         Rk 
                         
                           Rj 
                           + 
                           Rk 
                         
                       
                     
                     ) 
                   
                 
               
               
                 
                   Equation 
                    
                   
                       
                   
                    
                   14 
                 
               
             
           
         
       
     
         [0000]    When the ambient temperature is held constant and the effects of self-heating are minimized (such that Rh does not change), resistance values in the bridge circuit will not change and the ratio p of the output voltage V out  to the gain G multiplied by the source-ground voltage difference (V cc −V G ) will be relatively constant. As shown in Equation 15, when ρ remains constant ΔV out /(G·ΔV cc ) is also constant: 
         [0000]    
       
         
           
             
               
                 
                   
                     
                       ∂ 
                       
                         V 
                         out 
                       
                     
                     
                       ∂ 
                       
                         V 
                         cc 
                       
                     
                   
                   = 
                   
                     G 
                     · 
                     ρ 
                   
                 
               
               
                 
                   Equation 
                    
                   
                       
                   
                    
                   15 
                 
               
             
           
         
       
     
         [0103]      FIG. 10   a  illustrates a flow chart describing a self-heating calibration process  1000   a  in accordance with another aspect of the present invention for identifying the voltage settings at which ΔV out /(G·ΔV cc ) is constant, and wherein this information is utilized to calibrate the temperature measurement circuit against self-heating. In some embodiments, the self-heating calibration process  1000   a  may be performed by a self-heating calibration controller. In one embodiment, this process involves setting the supply voltage V cc  at a desired voltage level V op  causing an unknown amount of self-heating and an unknown temperature change ΔT op . The supply voltage V cc  is then gradually lowered, which reduces the effect of self-heating. Because the temperature change ΔT is proportional to the square of the current, the effects of self-heating will rapidly diminish as the voltage V cc  is reduced. At a sufficiently low supply voltage V L , the effects of self-heating will be negligible. As described above, the low supply voltage V L  at which self-heating effects are no longer observed can be identified as the point at which the ratio ρ of the output voltage V out  to the gain G multiplied by the supply-source voltage difference (V cc −V G ) remains relatively constant despite changes to the supply voltage V cc . At this point, a comparison between the ratio ρ op  at the desired voltage level V op  with the ratio ρ L  at the low voltage level V L  may be used to determine the effect ΔV sh  of self-heating at the desired source voltage level using the relation shown in Equation 16: 
         [0000]      Δ V   sh   =G ·(ρ op   −ρ   L )·( V   op     31  V   G )   Equation 16 
         [0104]    As shown in  FIG. 10   a,  the self-heating calibration process  1000   a  may begin at step  1002 , where the self-heating calibration controller sets the source voltage V cc  to be equal to a desired operational voltage V op  such as, for example, the voltage that will be desired to provide a good common mode signal and the ability to rapidly heat the RTD. 
         [0105]    At step  1003 , the self-heating calibration controller sets the gain G for the programmable gain instrumentation amplifier  302  to a desired operational gain value G op  such as, for example, the gain value that will maximize the signal-to-noise ratio of the temperature measurement circuit  300 . 
         [0106]    After these initial values are set, at step  1004 , the self-heating calibration controller measures the output voltage V out  of the temperature measurement circuit  300 . 
         [0107]    At step  1005 , the self-heating calibration controller calculates a ratio Ratio op  between the measured output voltage V out  and the operational supply-ground voltage (V cc −V G ) multiplied by the operational gain G op . 
         [0108]    After the initial ratio Ratio op  is calculated, at step  1006 , the self-heating calibration controller decreases the source voltage V cc . After the source voltage V cc  has been decreased, the self-heating calibration controller measures the new output voltage V out  at step  1007  and calculates a new ratio Ratio i  at step  1008 . 
         [0109]    At step  1009 , the self-heating calibration controller determines the difference between the newly calculated ratio Ratio i  and the previously calculated ratio Ratio i-1 . In the case that the difference between these ratios is above a predetermined threshold, that is, decreasing the source voltage V cc  continues to have a substantial effect on the ratio, the self-heating calibration controller will return to step  1006  and lower the source voltage V cc  again. In the case that the difference between these ratios is below the threshold (that is, decreasing the source voltage V cc  no longer has a substantial effect on the ratio), then the self-heating calibration controller has identified V L  and will proceed to step  1011 . 
         [0110]    At step  1011 , the self-heating calibration controller calculates the self-heating voltage difference ΔV sh  by subtracting the ratio Ratio i  calculated at the low voltage V L  from the ratio Ratio op  calculated at the operational voltage V op  and multiplying the difference by the gain of the programmable gain instrumentation amplifier  302  and the desired operational source-ground voltage, in accordance with Equation 16. 
         [0111]    Referring to  FIG. 11 , a graph of the ratio ρ vs. the source voltage V cc  for an exemplary embodiment of the invention is illustrated. As can be seen in  FIG. 11 , at relatively high source voltages (e.g. V cc =30 v), the slope of the graph is relatively higher, that is, ρ is not constant. At relatively low source voltages (e.g. V cc =10 v), the slope of the graph is substantially lower. At source voltages below 10 volts, the value of the ratio p approaches a horizontal asymptote, that is, ρ remains relatively constant. To calculate ΔV sh , the value of ρ at the asymptote is subtracted from the value of ρ at the desired operational configuration. 
         [0112]      FIG. 10   b  illustrates a flow chart describing a self-heating calibration process  1000   b  in accordance with another embodiment of the present invention. As illustrated in  FIG. 10   b,  the self-heating process  1000   b  is similar to self-heating process  1000   a.  In this embodiment, at steps  1004   b  and  1007   b,  the self-heating calibration controller measures the source voltage V cc  using the same voltage measurement system as that used to measure V out . This provides a real time measurement of the supply voltage V cc  that can be used for the normalization self-heating calibration process and may provide a more accurate value for V cc  than simply relying on an input voltage setting. 
         [0113]    Referring to  FIG. 10   c,  a flow chart describing a self-heating calibration process  1000   c  in accordance with another embodiment is illustrated. As illustrated in  FIG. 10   c,  the self-heating process  1000   c  is similar to self-heating process  1000   a.  In this embodiment, at step  1006   b - 2 , the self-heating calibration controller increases the gain G of the programmable gain instrumentation amplifier  302 . This increase, combined with the decrease of the source-ground voltage (V cc −V G ), may be used to ensure a sufficient range of output voltages V out  even when the source-ground voltage (V cc −V G ) is small. In some embodiments, the gain G and the source voltage V cc  may be varied according to an inverse relationship such that the product of the source-ground voltage (V cc −V G ) and the gain G remains constant. For example, suppose the gain G is initially 10 and the source-ground voltage (V cc −V G ) is 30 v. A decrease in the source voltage V cc  of 5 volts (so the source-ground voltage is now 25 volts) should be accompanied by an increase in the gain G from 10 to 12. 
         [0114]    Thermal Calibration 
         [0115]    With the chip  102 ,  202  loaded and the gross adjustments made to account for changes in heater resistance and changes in factors associated with chip loading, a fine thermal calibration may be desirable. Because PCR efficacy and diagnosis based on thermal melt depend heavily on the accuracy of the temperature measurement, thermal calibration may be required immediately before a diagnostic cycle begins. By measuring the voltage response versus temperature or the implied resistance versus temperature relationship (where resistance may, for example, be determined based on Equation 4 above) the system can define a precise calibration for the RTD immediately before a microfluidic chip  102 ,  202  is used. 
         [0116]      FIG. 12  illustrates a flow chart describing a thermal calibration process  1200  in accordance with another aspect of this invention. In some embodiments, the thermal calibration process  1200  may be performed by a thermal calibration controller. As shown in  FIG. 12 , process  1200  may begin at step  1202 , where the thermal calibration controller may set the source voltage V cc  to the desired operational voltage V op . 
         [0117]    At step  1203 , the thermal calibration controller may set the gain value G for the programmable gain instrumentation amplifier  302  to a desired operational gain voltage G op . The values of V op  and G op  may be selected, for example, to match the values that will be used when the microfluidic device  202  is performing PCR or high-resolution melt. 
         [0118]    After the initial values have been set, the thermal calibration controller brings the temperature detecting circuit  300  to a predetermined temperature T n  at step  1204 . In some embodiments, this is achieved by utilizing an externally controlled heating device (e.g. a Peltier device, a resistive heater, etc.). 
         [0119]    At step  1205 , the thermal calibration controller measures the output voltage V n . At step  1206 , the thermal calibration controller stores the values of the temperature T n  and the output voltage V n  are stored in association with each other. 
         [0120]    Then, the thermal calibration controller  1200  returns to step  1204  wherein the temperature detecting circuit  300  is brought to a new temperature T n . This is repeated until the thermal calibration controller measures and stores a predetermined number M of (T n , V n ) relationships. 
         [0121]    With the stored set of (T n , V n ), the temperature control system  120  can determine precise values for temperature detection. The interpolation may take the form of a curve with one or more constants for each resistive sensor on the platform chip (such as a 3 term quadratic calibration curves), or calibration may take the form of a look-up table with set voltages (or resistances) for each temperature. 
         [0122]    In some embodiments, the externally controlled heating device is able to generate a uniform temperature environment for the platform chip and is able to precisely measure temperature. The external temperature measurement may be made by any suitable device including an RTD, a thermocouple, a thermistor, a semiconductor junction device, etc. The external temperature measurement device should be factory or third party calibrated and its calibration data should be embedded in control software, which may be configured to include this calibration data as part of thermal calibration process  1200 . 
         [0123]    Embodiments of the present invention have been fully described above with reference to the drawing figures. Although the invention has been described based upon these preferred embodiments, it would be apparent to those of skill in the art that certain modifications, variations, and alternative constructions could be made to the described embodiments within the spirit and scope of the invention. 
         [0124]    Additionally, while the process described above and illustrated in the drawings is shown as a sequence of steps, this was done solely for the sake of illustration. Accordingly, it is contemplated that some steps may be added, some steps may be omitted, the order of the steps may be re-arranged, and some steps may be performed in parallel.