Abstract:
A capacitive sensing circuit is disclosed, wherein the mixer is directly connected to the sense electrode. The AC transimpedance amplifier in front of the mixer in prior art is removed and replaced by a differential DC transimpedance amplifier respectively integrator. The mixer DC offset voltage or current together with the large amplification factor required after the mixer now would result in an inacceptable DC offset at the output of the signal chain. In order to eliminate the effect of the mixer offset, the amplifying stages after the mixer are AC coupled to the mixer output and one of the signals entering the mixer is phase modulated or amplitude modu

Description:
FIELD OF THE INVENTION 
       [0001]    The present invention generally relates to the technical field of capacitive measurement circuits and more specifically to a capacitive measurement system having one or more electrodes, in which the characteristics of a conductive body such as shape and location are determined by means of capacitive coupling via the electrically conductive body. 
       BACKGROUND OF THE INVENTION 
       [0002]    Capacitive measurement and/or detection systems have a wide range of applications, and are among others widely used for the detection of the presence and/or the position of conductive body in the vicinity of an electrode of the system. A capacitive sensor, called by some electric field sensor or proximity sensor, designates a sensor, which generates a signal responsive to the influence of what is being sensed (a person, a part of a person&#39;s body, a pet, an object, etc.) upon an electric field. A capacitive sensor generally comprises at least one antenna electrode, to which is applied an oscillating electric signal and which thereupon emits an electric field into a region of space proximate to the antenna electrode, while the sensor is operating. The sensor comprises at least one sensing electrode—which could comprise the one or more antenna electrodes themselves—at which the influence of an object or living being on the electric field is detected. 
         [0003]    The technical paper entitled “Electric Field Sensing for Graphical Interfaces” by J. R. Smith, published in Computer Graphics I/O Devices, Issue May/June 1998, pp 54-60 describes the concept of electric field sensing as used for making non-contact three-dimensional position measurements, and more particularly for sensing the position of a human hand for purposes of providing three dimensional positional inputs to a computer. Within the general concept of capacitive sensing, the author distinguishes between distinct mechanisms he refers to as “loading mode”, “shunt mode”, and “transmit mode” which correspond to various possible electric current pathways. In the “loading mode”, an oscillating voltage signal is applied to a transmit electrode, which builds up an oscillating electric field to ground. The object to be sensed modifies the capacitance between the transmit electrode and ground. In the “shunt mode”, which is alternatively referred to as “coupling mode”, an oscillating voltage signal is applied to the transmit electrode, building up an electric field to a receive electrode, and the displacement current induced at the receive electrode is measured, whereby the displacement current may be modified by the body being sensed. In the “transmit mode”, the transmit electrode is put in contact with the user&#39;s body, which then becomes a transmitter relative to a receiver, either by direct electrical connection or via capacitive coupling. 
         [0004]    The capacitive coupling is generally determined by applying an alternative voltage signal to a capacitive antenna electrode and by measuring the current flowing from said antenna electrode either towards ground (in the loading mode) or into the second electrode (receiving electrode) in case of the coupling mode. This current is usually measured by means of a transimpedance amplifier, which is connected to the sensing electrode and which converts a current flowing into said sensing electrode into a voltage, which is proportional to the current flowing into the electrode. 
         [0005]      FIG. 1  shows a typical prior art circuit configured to measure an unknown capacitance in so-called ‘loading’ mode meaning that the capacitance between an electrode of a capacitive sensor and ground or earth is measured. 
         [0006]    An AC voltage source  1  generates an AC voltage signal of known frequency and amplitude, for example a periodic sine wave of 100 kHz and 1 V peak amplitude. The output node  2  of AC voltage source  1  is connected to the non-inverting input of an operational amplifier  3 . Operational amplifier  3  is configured as transimpedance amplifier. Operational amplifier  3 , through the feedback action of associated feedback impedance  4  (preferably a capacitance connected in parallel to a resistance, whereby the impedance of the capacitance at the operating frequency is at least 10 times smaller than the resistance), maintains substantially the same potential on its inverting input as on its non-inverting input, thereby keeping sense node  5  at the same potential than AC voltage source output  2 . The unknown capacitance  6  to be measured accordingly has the AC voltage source voltage applied across its “plates”. 
         [0007]    The current flowing through unknown capacitance  6  is then given by its capacitance and the known AC voltage source voltage, said current flowing also through feedback impedance  4  as the input current into the non-inverting input of amplifier  3  is substantially zero. 
         [0008]    The voltage on output  7  of amplifier  3  is accordingly responsive to the AC voltage source voltage and the unknown capacitance. This amplifier output voltage is then mixed with mixer  8  (for example a switching mixer or a multiplier) whereby the local oscillator input of mixer  8  is driven by the AC voltage source output  2 . The output of mixer  8  is a DC voltage superimposed with multiples of the AC voltage source frequency, the DC voltage level being responsive to the amplitude of the amplifier output  7  and thereby of AC voltage source output voltage  2  and unknown capacitance  6 . 
         [0009]    As only the DC voltage is desired, the multiples of the AC voltage source frequency are filtered out with low pass filter  10 . The output signal  11  of the low pass filter is a DC voltage responsive to the AC voltage source voltage and the unknown capacitance. Furthermore, an adjustable phase shift (preferably of selectable steps of 0 and 90 degrees) can be introduced between the AC voltage source output  2  and local oscillator input of mixer  8 , thereby allowing the measurement of the complex impedance  6  instead of a capacitance  6 . 
         [0010]      FIG. 2  shows a typical prior art circuit configured to measure an unknown capacitance in so-called ‘coupling’ mode meaning that the capacitance between two electrodes of a capacitive sensor is measured. 
         [0011]    In this variant, an AC voltage source  1  generates an AC voltage signal of known frequency and amplitude, for example a periodic sine wave of 100 kHz and 1 V peak amplitude. The output node  2  of AC voltage source  1  is connected to the first plate of unknown capacitance  6 . The second plate of unknown capacitance  6  is connected to the inverting input of an operational amplifier  3 . The non-inverting input of amplifier  3  is connected to ground. Operational amplifier  3 , through the feedback action of the associated feedback impedance  4  (preferably a capacitance connected in parallel to a resistance, whereby the impedance of the capacitance at the operating frequency is at least 10 times smaller than the resistance), maintains substantially the same potential on its inverting input as on its non-inverting input, thereby keeping sense node  5  at ground potential. The unknown capacitance  6  to be measured accordingly has the AC voltage source voltage applied across its “plates”. 
         [0012]    The current flowing through unknown capacitance  6  is then given by its capacitance and the known AC voltage source voltage, said current flowing also through feedback impedance  4  as the input current into the non-inverting input of amplifier  3  is substantially zero. 
         [0013]    The voltage on output  7  of amplifier  3  is accordingly responsive to the AC voltage source voltage and the unknown capacitance. This amplifier output voltage is then mixed with mixer  8  (for example a switching mixer or a multiplier) whereby the local oscillator input of mixer  8  is driven by the AC voltage source output  2 . The output of mixer  8  is a DC voltage superimposed with multiples of the AC voltage source frequency, the DC voltage level being responsive to the amplitude of the amplifier output  7  and thereby of AC voltage source output voltage  2  and unknown capacitance  6 . 
         [0014]    As only the DC voltage is desired, the multiples of the AC voltage source frequency are filtered out with low pass filter  10 . The output signal  11  of the low pass filter is the DC voltage responsive to the AC voltage source voltage and the unknown capacitance. Furthermore, an adjustable phase shift (preferably of selectable steps of 0 and 90 degrees) can be introduced between the AC voltage source output  2  and local oscillator input of mixer  8 , thereby allowing the measurement of the complex impedance  6  instead of a capacitance  6 . 
         [0015]    For both prior art circuits, the gain of the transimpedance amplifier formed by the operational amplifier  3  and the feedback impedance  4  is configured to be as large as possible in order to achieve low noise performance, and the DC gain of the signal chain stages following the mixer can subsequently be made comparatively low, to avoid DC offset problems. For example, in a practical implementation, for an operating frequency of 100 kHz and a source amplitude of 1 V, the feedback impedance would be chosen to be a capacitor of 100 pF in parallel with a resistance of 1 MΩ. 
         [0016]    However, the output signal range of the operational amplifier  3  is limited, for example to an amplitude of 2 V peak for a 5 V power supply. This implies that a parasitic AC current injected into the sense electrode of the capacitive sensor of more than 126 μA peak amplitude will drive the operational amplifier into saturation and introduce an error into the measurement of the unknown capacitance. Such parasitic AC currents are e.g. generated by external noise sources, one example being the so-called ‘Bulk current injection’ (BCI) test during the qualification of an occupant detection system. 
       OBJECT OF THE INVENTION 
       [0017]    The object of the present invention is to provide a robust capacitive measurement circuit, which is less sensitive to such parasitic AC currents. 
       GENERAL DESCRIPTION OF THE INVENTION 
       [0018]    In order to overcome the abovementioned problems, the present invention proposes a capacitive sensing circuit, wherein the mixer is connected upstream of the amplifying stage. The AC transimpedance amplifier upstream of the mixer in the prior art circuits is removed and replaced by a differential DC transimpedance amplifier respectively by an integrator. 
         [0019]    The mixer DC offset voltage or current together with the large amplification factor required after the mixer now would result in an inacceptable DC offset at the output of the signal chain. In order to eliminate the effect of the mixer offset, the amplifying stages after the mixer are AC coupled to the mixer output and one of the signals entering the mixer is phase modulated or amplitude modulated with a known low frequency signal. An additional mixer after the AC coupled amplifying stages is driven with the same low frequency modulating signal, resulting in the wanted DC output signal responsive to the capacitance to be measured. 
         [0020]    In a first preferred embodiment the capacitive detection system comprises an antenna electrode, a first AC signal generator configured to generate a first AC voltage signal, a second AC signal generator configured to generate a second AC voltage signal, said second AC voltage having a lower frequency than said first AC signal, and a first mixer for mixing said first AC voltage signal and said second AC voltage signal and for generating a modulated AC voltage signal. One of said first AC signal generator or said first mixer is operatively coupled to said antenna electrode to apply said first AC voltage signal or said modulated AC voltage signal to said antenna electrode. 
         [0021]    The capacitive detection system further comprises a control and evaluation unit operatively coupled to said antenna electrode or a separate receiving electrode, said control and evaluation unit comprising a current measurement circuit configured to measure current signals, said current signals comprising amplitude and/or phase of a current flowing in said antenna electrode or in said separate receiving electrode, said control and evaluation unit being configured to determine a capacitance to be measured based upon said measured current signals, and to output an output signal indicative of said determined capacitance. According to one aspect of the invention said current measurement circuit comprises a differential transimpedance amplifier circuit comprising a common mode voltage setting input, an inverting input, a non-inverting input and an output, wherein said inverting input and said non-inverting input of said transimpedance amplifier circuit are operatively connected to said antenna electrode or said separate receiving electrode by means of a multiplexer in such a way that said inverting input and said non-inverting input are alternately supplied with said current flowing in said antenna electrode or in said separate receiving electrode, said multiplexer being operatively coupled to and controlled by the other one of said first AC signal generator or said first mixer. 
         [0022]    In one variant of the above system, said first AC signal generator is operatively coupled to said antenna electrode to apply said first AC voltage signal to said antenna electrode and said multiplexer is operatively coupled to said first mixer and controlled by said modulated AC voltage signal. In an alternative variant said first mixer is operatively coupled to said antenna electrode to apply said modulated AC voltage signal to said antenna electrode and said multiplexer is operatively coupled to said first AC signal generator and controlled by said first AC voltage signal. 
         [0023]    In one embodiment of the capacitive detection system, said multiplexer is operatively coupled to the other one of said first AC signal generator or said first mixer by means of an adjustable phase shifter, so that said multiplexer is controllable by different phase positions of said first AC voltage signal or said modulated AC voltage signal. 
         [0024]    Finally an output signal at the output of said differential transimpedance amplifier circuit is preferably filtered by means of a bandpass filter and subsequently mixed with the second AC voltage signal of said second AC signal generator. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0025]    Further details and advantages of the present invention will be apparent from the following detailed description of not limiting embodiments with reference to the attached drawings, wherein: 
           [0026]      FIG. 1  shows a prior art ‘loading’ mode sensing circuit; 
           [0027]      FIG. 2  shows a prior art ‘coupling’ mode sensing circuit; 
           [0028]      FIG. 3  a first embodiment of a ‘loading’ mode sensing circuit according to the present invention; 
           [0029]      FIG. 4  an alternative embodiment of a ‘loading’ mode sensing circuit according to the present invention; 
           [0030]      FIG. 5  a first embodiment of a ‘coupling’ mode sensing circuit according to the present invention; 
           [0031]      FIG. 6  an alternative embodiment of a ‘coupling’ mode sensing circuit according to the present invention; 
           [0032]      FIG. 7  shows a preferred embodiment of the circuit in  FIG. 3 . 
       
    
    
     DESCRIPTION OF PREFERRED EMBODIMENTS 
       [0033]    The circuit shown in  FIG. 3  is a first embodiment allowing to substantially improve the immunity of the capacitance measurement circuit against injection of external parasitic AC currents. The capacitive measurement circuit is working in the so-called loading mode. 
         [0034]    AC voltage source  21  generates an AC voltage signal of known frequency and amplitude, for example a periodic sine wave of 100 kHz and 1 V peak amplitude. Its output node  22  is connected to the input of adjustable phase shifter  32 , and a first input of mixer  23 . A second AC voltage source  24  generates a second AC voltage signal of know frequency and amplitude, but of lower frequency than the output frequency of AC voltage source  21 , for example a periodic square wave of 1 kHz and 1V peak amplitude. 
         [0035]    The output  25  of AC voltage source  24  is connected to the second input, the local oscillator input, of mixer  23 . Mixer  23  multiplies the signals at its two inputs. For the specific example signals described above, a phase modulated sine wave will be generated at its output  26 , that is for the first half of the period of the output signal of AC voltage source  24 , the output  26  will be identical to the output signal of AC voltage source  21 , and during the second half of the period of AC voltage source  24 , the output  26  will be the inverted version of the output signal of AC voltage source  21 . 
         [0036]    Obviously, different waveforms can be used instead of the square waveform for AC voltage source  24 , for example a so-called pseudo random noise binary sequence, or a swept frequency or stepped frequency square wave. 
         [0037]    The output  26  of mixer  23  is fed into the sense to guard capacitance  43  of the capacitive sensor, and into the common mode voltage setting input of differential transimpedance amplifier  40 . The differential transimpedance amplifier  40  keeps both of its inputs  44  and  45  at the same AC voltage level than its common mode voltage setting input, therefore nodes  44  and  45  are at the same AC potential than node  26 . As multiplexer  30  always switches one of the left plates of capacitors  41  or  42  to the sense node  29 , the sense node  29  thereby also has always the same AC voltage than node  26 . Therefore, the AC voltage across sense to guard capacitance  43  is substantially zero, which allows the use of a guard electrode connected to node  26  to shield the sense electrode connected to node  29  against unwanted parasitic capacitances between sense node  29  and ground. 
         [0038]    Also, unknown capacitance or impedance  28  has also the known AC voltage of node  26  across its plates. The current flowing through unknown capacitance or impedance  28  is therefore determined by its impedance and said known AC voltage. Said current is also flowing through multiplexer  30 . The switch position of multiplexer  30  is steered by the polarity of the output signal of phase shifter  32 . Said current is therefore flowing through capacitor  41  into the positive input  44  of differential transimpedance amplifier  40 , or through capacitor  42  into the negative input  45  of differential transimpedance amplifier  40 . 
         [0039]    The differential transimpedance amplifier  40  amplifies the difference between the currents at its positive and negative inputs, and outputs an output voltage responsive to the input current difference. Multiplexer  30 , together with differential transimpedance amplifier  40  therefore constitute a switching synchronous rectifier or switching synchronous demodulator with a current mode input and low AC input impedance, as the AC voltage on node  29  is substantially equal to the voltage on node  26  and does substantially not depend on the unknown capacitance or impedance  28 . 
         [0040]    The DC output voltage of said synchronous rectifier is responsive to the known AC voltage on node  29 , the unknown capacitance or impedance  28  and the phase shift adjusted with the phase shifter  32 . Typically, the phase shift of phase shifter  32  is first set to 0 degrees, then a first measurement is performed, then the phase shift is set to 90 degrees, then a second measurement is performed. By doing two measurements, the complex impedance of the unknown capacitor respectively impedance  28  can be calculated. 
         [0041]    At the output  31  of differential transimpedance amplifier  40 , a first AC signal with the same frequency as the frequency of the AC voltage source  24  will appear, superimposed with a second image AC signal of the signal of AC voltage source  24 , shifted to twice the frequency of the output signal of AC voltage source  21 . Further images will also be produced at the harmonics of the output signal of AC voltage  21 . 
         [0042]    As only the first, low frequency AC signal is of interest for the capacitive measurement, the higher frequency components will be eliminated by the amplifier  33 , configured as bandpass filter, amplifying the first, low frequency AC signal and at the same time eliminating any DC offset signal at the output of mixer  31 , and at the same time, substantially suppressing any signal which has frequency components other than the wanted, first low frequency signal. Amplifier  33  can for example be configured for the assumed 1 kHz output frequency of AC voltage source  24 , with an AC coupled (capacitive coupled) 4-pole Butterworth lowpass filter with a cutoff frequency of 1.5 kHz, implemented for example with two operational amplifiers in the Sallen-Key configuration. 
         [0043]    The resulting 1 kHz signal  34  at the output of bandpass amplifier  33  is then again mixed with the AC output signal of AC signal source  24  by mixer  35 , and then amplified and low-pass filtered by amplifier  37  configured as low pass filter. Amplifier  37  can for example be implemented with a DC-coupled 2-pole Butterworth lowpass filter with a 100 Hz cutoff frequency, implemented for example with one operational amplifiers in the Sallen-Key configuration. 
         [0044]    Another preferable, less complex option is to replace amplifier  37  with a passive low pass RC filter, having a DC gain of one when the amplifier  33  has been chosen to have sufficient gain for the application. 
         [0045]    The DC voltage at the final output  38 , will then due to the action of mixer  35  and low-pass action of amplifier  37 , be responsive to the amplitude of the 1 kHz signal at the input of mixer  34 . Finally, said DC voltage is responsive to the current through the unknown capacitance respectively impedance  28 . 
         [0046]    By performing the two consecutive measurements described above (the first one with phase shifter  32  adjusted to 0 degrees phase, the second one with phase shifter  32  adjusted to 90 degrees phase shift), and by combining the two consecutive DC levels obtained at the output  38 , the impedance of the unknown capacitance respectively impedance  28  can be calculated. 
         [0047]    The sequencing of the measurements and the measurement of the DC level at output  38  and the computation of the impedance of the unknown capacitance respectively impedance  28 , is preferably performed by a microcontroller equipped with an integrated ADC (analog to digital converter). Alternatively, mixer  35  and low-pass filtering amplifier  37  can all be implemented inside a microcontroller equipped with an ADC, by connecting the ADC input directly to the output  34  of amplifier  33 , and implementing the mixer in software by multiplying the ADC results alternatively with the values +1 and −1, synchronized to the AC voltage source  24 , and then low-pass filtering or integrating the resulting values by software. 
         [0048]    The differential transimpedance amplifier  40  can also be interpreted as a differential current mode input integrator, whereby the integrator keeps the AC voltages of each of the inputs  44  and  45  at the same AC potential than its common mode voltage setting input. 
         [0049]    In order to optimally suppress injected parasitic AC currents, it is preferable to first perform a sweep or stepped scan of the frequency of the AC voltage source  21 , detect the frequency or frequencies where the parasitic AC currents are located, and then set the measurement frequency of AC voltage source  21  to a frequency at which no parasitic AC current has been detected, and where also no subharmonic of a parasitic AC current is present. 
         [0050]    An alternative to the circuit shown in  FIG. 3  is the circuit shown in  FIG. 4 . The capacitive measurement circuit is working in the so-called loading mode. The difference to the circuit in  FIG. 3  is that the common mode voltage setting input of differential transimpedance amplifier  40  and the guard electrode (top plate of guard to sense capacitor  43 ) are now directly connected to the output  22  of the AC voltage source  21  and the input of phase shifter is connected to the output of mixer  26 . The rest of the operation of the circuit is identical to the circuit in  FIG. 3 , except that common mode voltage setting input of differential transimpedance amplifier  40  and unknown capacitance respectively impedance  28  is now fed with a continuous wave, non modulated, periodic signal instead of a modulated one as in  FIG. 3 . 
         [0051]      FIG. 5  shows a circuit similar to the circuit in  FIG. 3 , except that the capacitive measurement circuit is working in the so-called coupling mode. Therefore, only the differences to the circuit in  FIG. 3  will be described. 
         [0052]    The output  26  of mixer  23  is connected to the unknown capacitance or impedance to be measured  28 . The other node of the capacitance or impedance to be measured  28  is connected to the multiplexer  30 . As the common mode voltage setting input of differential transimpedance amplifier  40  is connected to ground, the AC voltage on node  29 , through capacitors  41  and  42  and multiplexer  30 , will have zero AC voltage. Therefore, the known AC voltage of node  26  will also be present across unknown capacitance or impedance  28 . 
         [0053]    Similar to the circuit in  FIG. 3 , the current through unknown capacitance or impedance  28  will then be defined by the known AC voltage of node  26 , and the impedance of unknown capacitance or impedance  28 . Furthermore, as in  FIG. 3 , the DC voltage of output  38  will be responsive to said impedance. 
         [0054]    An alternative to the circuit shown in  FIG. 5  is the circuit shown in  FIG. 6 . The capacitive measurement circuit is working in the so-called coupling mode. The difference to the circuit in  FIG. 5  is that the unknown capacitance or impedance to be measured  28  is now directly connected to the output  22  of the AC voltage source  21  and the input of phase shifter is connected to the output of mixer  26 . The rest of the operation of the circuit is identical to the circuit in  FIG. 5 , except that the unknown capacitance respectively impedance  28  is now fed with a continuous wave, non modulated, periodic signal instead of a modulated one as in  FIG. 5 . 
         [0055]      FIG. 7  shows one preferred embodiment of the circuit in  FIG. 3 . Only the differences of the circuit in  FIG. 7  compared to the circuit in  FIG. 3  will be described. The differential transimpedance amplifier  40  in  FIG. 3  is replaced by operational amplifiers  411  and  421 , capacitors  413  and  423  and resistors  414  and  424  and difference amplifier  430 . Difference amplifier  430  is preferably implemented with a suitably configured operational amplifier, an example can be found in reference “The Art of Electronics, 2 nd  edition, Paul Horowitz and Winfield Hill”, page 185, FIG. 4.18. 
         [0056]    Operational amplifiers  411  and  412  are configured identically, so only the configuration of operational amplifier  411  will be described. Capacitor  41  couples the AC current coming from multiplexer  30  into the transimpedance amplifier made out of operational amplifier  411  and associated components. Capacitor  413  and resistor  414  close the feedback path around the operational amplifier and determine the gain of the transimpedance amplifier. Through the action of the feedback components  413  and  414 , the voltage difference between the inputs of operational amplifier  411  is substantially kept to zero volts. As the positive input is connected to the output  26  of mixer  23 , the input of the transimpedance amplifier built around operational amplifier  411  connected to capacitor  41  is at the same said AC potential, and, as operational amplifier  421  is configured identically, the input of the transimpedance amplifier built around operational amplifier  421  connected to capacitor  42  is also at the same said AC potential. Preferred values for capacitors  41  and  42  are 500 nF, for capacitors  413  and  423  10 nF, and for resistors  414  and  424  500 kΩ. 
         [0057]    An example for the operational amplifiers  411  and  421  is the LT1057 from Linear Technology. The voltage gain of the difference amplifier is preferably set to 10. As there is no amplifying device between the input of the capacitive measurement circuit and the mixer, and as the differential transimpedance amplifier is only required to amplify DC signals, the capacitive measurement circuit is substantially more insensitive to parasitic AC current injected into the sense electrode of the capacitive sensor (node  29  in  FIG. 7 ). For example, for the circuit in  FIG. 7  with the components as defined above, an injected parasitic current of 10 mA peak amplitude does not notably falsify the measurement result, which compares favourably to the 126 μA peak amplitude for the prior art circuit in  FIG. 1 .