Abstract:
The invention addresses itself to the task of providing a medium-frequency stereo broadcast receiving circuit that does not require any alteration of the radio wave format used in the medium-frequency stereo broadcast system that has become the de facto standard, and that receives transmitted broadcast waves and removes, in the course of the demodulation process, disturbance that has affected the signal during its propagation, thereby improving the audio quality of the demodulated signal and obtaining the full potential of the stereo effect. To achieve this, the received medium-frequency stereo broadcast wave is converted to a single-sideband signal, and the sum signal (L+R) is demodulated from the phase term of this converted single-sideband signal. The difference signal (L−R) is demodulated from the phase term of the received medium-frequency stereo broadcast wave and from the demodulated sum signal output.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    The present invention relates to a receiving circuit for receiving medium-frequency stereo broadcasts. Compatible-Quadrature amplitude modulation (C-Quam), a system devised by the Motorola Corporation of the USA, has become the de facto standard for medium-frequency stereo broadcasting. The present invention relates to a technique for improving the audio quality of the demodulated signal while obtaining a full stereo effect in a medium-frequency stereo broadcast receiving circuit for receiving and demodulating a C-Quam broadcast wave.  
           [0003]    2. Description of Related Art  
           [0004]    Medium-frequency stereo broadcasting first began in the USA in 1982. It was implemented in Australia in 1985, in Brazil in 1986, in Canada in 1988, and in Japan in 1992. In the USA, four systems have been proposed in addition to C-Quam, but these other systems have been suppressed and at present C-Quam is the main system in use, with over 600 stations already using it for stereo broadcasts. Numerous other countries including Japan have adopted the C-Quam system as the standard system, and outside the USA a total of over 150 stations are now offering C-Quam based stereo broadcasts.  
           [0005]    In order to maintain compatibility with conventional medium-frequency broadcast receivers, C-Quam forms a sum signal (L+R) and a difference signal, (L−R) from the left and right information signals, forms an angle-modulated wave that has been modulated by these sum and difference signals, and transmits a signal obtained by amplitude modulating this angle-modulated wave with the sum signal (L+R). Because the component that has been amplitude-modulated by the sum signal (L+R) can be received by a conventional monophonic medium-frequency receiver, compatibility is assured.  
           [0006]    However, the following problems have been encountered with conventional C-Quam demodulation technology: 
           [0007]    1. Because the C-Quam system uses quadrature detection with synchronous detection of the in-phase modulation component (the I-channel) and the quadrature-modulation component (the Q-channel), accurate tuning is required to demodulate the sum signal (L+R) and the difference signal (L−R).  
           [0008]    2. In order to obtain a stereo signal using the C-Quam system, the difference signal (L−R) has to be demodulated from the quadrature-modulation component (the Q-channel) by means of synchronous detection, and this necessitates processing that accurately removes redundant modulation components contained in the received signal. However, because it is difficult to perform this processing accurately, the audio quality of the demodulated signal during stereo reception is poorer than the audio quality during reception of a monophonic broadcast.  
           [0009]    3. A shortcoming of the conventional demodulation method employed in C-Quam is that because it utilizes the amplitude component of the broadcast wave, it is susceptible to external noise. In an actual receiver, this susceptibility is eliminated as much as possible by narrowing the pass band of the main band-limiting filter. However, this makes it difficult to obtain a high-fidelity demodulated signal from the transmitted information signal.  
           [0010]    4. Because these problem are encountered with the C-Quam system, existing receivers do not display the stereo effect to its full potential and cannot secure truly excellent audio quality.  
         SUMMARY OF THE INVENTION  
         [0011]    It is an object of the present invention to provide a medium-frequency stereo broadcast receiving circuit that overcomes the above-mentioned problems; that does not alter the radio wave format used in the medium-frequency stereo broadcast system that has become the de facto standard; and that receives transmitted broadcast waves and removes, in the course of the demodulation process, disturbance that has affected the signal during its propagation, thereby improving the audio quality of the demodulated signal and obtaining the full potential of the stereo effect.  
           [0012]    The medium-frequency stereo broadcast receiving circuit of this invention receives and demodulates a medium-frequency stereo broadcast wave—and in particular, a C-Quam broadcast wave—comprising an angle-modulated wave that has been modulated by the sum signal (L+R) and by the difference signal (L−R) of the left and right information signals, and which has also been amplitude-modulated by the sum signal. This medium-frequency stereo broadcast receiving circuit comprises: sum signal demodulation means for converting the received medium-frequency stereo broadcast wave to a single-sideband signal containing a carrier, and for demodulating the sum signal from the phase term of this converted single-sideband signal; and difference signal demodulation means for demodulating the difference signal from the phase term of the received medium-frequency stereo broadcast wave and the demodulated output of the sum signal demodulation means.  
           [0013]    The present invention introduces the inventive step of demodulating both the sum and the difference signals from the phase term of the C-Quam modulated signal. The reason for doing this is that the information signal component present in the phase term of the modulated signal is not readily susceptible to the effect of multiplicative or additive external noise, and as a result exhibits excellent transmission quality. The superior reception characteristics of an FM broadcast wave compared with an AM broadcast wave are likewise due to the fact that the information signal component in a frequency modulated signal is present only in the phase term and is demodulated from this phase term.  
           [0014]    We have therefore employed a demodulation processing method which removes the modulation component contained in the phase term of the amplitude-modulated signal, converts this modulation component to an RZ SSB signal, and then demodulates the sum signal (L+R) from the phase term of this RZ SSB signal. A demodulation processing technique of this sort is known as Real Zero Single Sideband (RZ SSB) modulation and demodulation, and is capable of removing, during the demodulation process, amplitude distortion due to external noise. Details of RZ SSB modulation and demodulation are given in JP H06-018333 B (granted as Japanese Patent No. 1888866).  
           [0015]    To remove the modulation component contained in the phase term of the amplitude-modulated signal, the sum signal demodulation means preferably comprises: first frequency conversion means for frequency converting the received medium-frequency stereo broadcast wave; means for branching the input signal to this first frequency conversion means and for limiting the amplitude of the branched portion of the signal; and second frequency conversion means for performing frequency conversion by multiplying together the output of this amplitude limiting means and the output of the first frequency conversion means.  
           [0016]    The signal that is required in order to extract the sum signal—i.e., a pure amplitude-modulated wave from which the modulation component due to the difference signal has been removed and which comprises only the signal component due to modulation by the sum signal—is obtained at the output of the second frequency conversion means. By converting this to a single-sideband signal, the sum signal can be extracted from the phase term of this signal, without concern that there are some extra signal components that have been overlooked. Moreover, the output of the second frequency conversion means is free of the influence of fading and frequency fluctuation.  
           [0017]    The second frequency conversion means and subsequent means are preferably provided in the intermediate frequency stage. This ensures that a high-quality demodulated signal is obtained irrespective of the frequency stability of the local oscillator in the high-frequency stage. As a result, the present invention does not forfeit the important feature of conventional envelope demodulation, namely, that demodulation characteristics are independent of frequency fluctuation. At the same time, it can accurately maintain the frequency characteristics of the transmitted information signal.  
           [0018]    The invention is also cleverly contrived so that the difference signal (L−R) as well can be demodulated from the phase term of the received signal. The method employed will be described below.  
           [0019]    A C-Quam medium-frequency stereo broadcast wave can be expressed as a function of time (t) by: 
             S ( t )=(1 +L+R ) cos (ω c   t +Φ( t )) 
           [0020]    where 
           tan Φ( t )=( L−R+P )/(1 +L+R ) 
           [0021]    and ω c  is the angular frequency of the carrier, (L+R) is the sum signal, (L−R) is the difference signal, and P is a pilot signal superimposed on the difference signal. Preferably, the difference signal demodulation means for demodulating the difference signal from a modulated signal of this sort comprises: a frequency discriminator for discriminating the frequency of the received medium-frequency stereo broadcast wave and extracting the angle component d/dt(Φ(t)); an integrator for integrating the extracted angle component d/dt(Φ(t)); a tangent function generator for generating the tangent function value tan Φ(t) of the output Φ(t) of this integrator; and means for multiplying together the output of this tangent function generator and the signal obtained by equalizing the delay of the output of the sum signal demodulation means and adding a suitable constant.  
           [0022]    An amplitude limiter (a hard limiter) is provided at the input to the difference signal demodulation means, but if amplitude limiting means is provided in the sum signal demodulation means, it would be possible to share this amplitude-limiting means by using it also as the amplitude limiter at the input to the difference signal demodulation means.  
           [0023]    Because an amplitude-modulated wave comprises an upper sideband and a lower sideband, the sum signal demodulation means preferably comprises frequency diversity means which, when converting the signal that has been amplitude-modulated by the sum signal (L+R) into a single-sideband signal, superimposes the received medium-frequency stereo broadcast wave and the signal obtained by reversing, in the frequency domain, the distribution of frequency components of this wave, and converts the resulting superimposed signal into one single-sideband signal.  
           [0024]    The frequency diversity means is provided in an intermediate frequency stage and can comprise: first frequency conversion means for multiplying together the medium-frequency stereo broadcast wave that has been converted to an intermediate frequency, and a local oscillator signal with a higher frequency than this carrier component, and for extracting the difference frequency component and the sum frequency component, which have mutually reversed distributions, in the frequency domain, of signal frequency components; means for branching the input signal to this first frequency conversion means and for limiting the amplitude of the branched portion of the signal; second frequency conversion means for (i) multiplying together the output of this amplitude limiting means and the difference frequency component extracted by the first frequency conversion means, and for extracting the sum frequency component, and for (ii) multiplying together the output of the amplitude limiting means and the sum frequency component extracted by the first frequency conversion means, and for extracting the difference frequency component; and means for adding the sum frequency component and the difference frequency component obtained by the second frequency conversion means.  
           [0025]    The medium-frequency stereo broadcast receiving circuit of this invention is preferably implemented using digital signal processing (DSP) technology, so that high-performance processing of the received signal can be carried out by an inexpensive circuit. Use of such technology renders circuit adjustment unnecessary and means that DSP processors can be used, which can be expected to offer volume production benefits. As a result, an economic receiver is assured. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0026]    Specific embodiments of the present invention will now be described, by way of example only, with reference to the accompanying of drawings in which:  
         [0027]    [0027]FIG. 1 is a block diagram of a medium-frequency stereo broadcast receiving circuit according to a first embodiment of the present invention;  
         [0028]    [0028]FIG. 2 is a block diagram of a medium-frequency stereo broadcast receiving circuit according to a second embodiment of the invention; and  
         [0029]    [0029]FIG. 3 is a block diagram of a medium-frequency stereo broadcast receiving circuit according to a third embodiment of the invention. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0030]    To aid an understanding of this invention, we give a brief description of a C-Quam transmitted wave. A wave transmitted after modulation according to the C-Quam scheme can be written as: 
           S ( t )=(1 +L+R ) cos (ω c   t +Φ(t))  (1) 
         [0031]    where 
         tan Φ( t )=( L−R+P )/(1 +L+R )  (2) 
         [0032]    and ω c  is the angular frequency of the carrier, (L+R) is the sum signal, (L−R) is the difference signal, and P is a pilot signal superimposed on the difference signal. To ensure that the amplitude-modulated component is not overmodulated, it is essential that: 
         | L+R|&lt; 1  (3) 
         [0033]    The following embodiments are described in terms of the use of a transmitted wave that can be expressed by Equation 1.  
         [0034]    First Embodiment  
         [0035]    A first specific embodiment of the present invention will now be described. FIG. 1 is a block diagram showing the configuration of this first embodiment, which comprises C-Quam transmitter  100 , transmitting antenna  101 , receiving antenna  102  of a C-Quam receiver, front-end amplifier  103 , frequency converter  104 , local oscillator  105 , intermediate frequency (IF) filter  106 , frequency converter  107 , local oscillator  108 , amplitude limiter (hard limiter)  109 , IF filter  110 , frequency converter  111 , IF filter  112 , RZ SSB demodulation processor  113 , delay circuit  114 , frequency discriminator  115 , band-pass filter  116 , integrator  117 , tangent function generator  118 , constant generator  119 , adder  120 , multiplier  121 , band-pass filter  122 , low-pass filter  123 , matrix circuit  124 , left audio signal output terminal  125 , right audio signal output terminal  126 , and pilot signal output terminal  127 .  
         [0036]    A brief description will now be given of signal flow in this first embodiment shown in FIG. 1, and of the functioning of its component circuits.  
         [0037]    The output of C-Quam transmitter  100  is transmitted as a C-Quam modulated wave by transmitting antenna  101 .  
         [0038]    The C-Quam modulated wave is received by antenna  102  of the C-Quam receiver, and after it has been amplified by front-end amplifier  103 , is converted by frequency converter  104  to an IF signal—for example, to the difference frequency between the received signal and the signal from local oscillator  105 . The required IF signal is then extracted by IF filter  106 .  
         [0039]    This extracted signal is split into two portions and one portion supplied to frequency converter  107  where the output of local oscillator  108  is used to convert it to the sum frequency signal. The required IF signal is then extracted by IF filter  110 . The other split portion of the signal is supplied to amplitude limiter (hard limiter)  109  where it is converted to a fixed-amplitude signal. The output of amplitude limiter (hard limiter)  109  is split into two portions and one portion supplied to frequency converter  111 , which functions so as to form the difference frequency signal between the output of frequency limiter  109  and the output of IF filter  110 . IF filter  112  extracts the lower sideband component of this difference frequency signal, this lower sideband component having had some unwanted noise components removed and being accompanied by a carrier component. The output of IF filter  112  is supplied to RZ SSB demodulation processor  113  which demodulates it, thereby obtaining the sum signal (L+R). This sum signal is supplied to delay circuit  114 .  
         [0040]    The other split portion of the output of amplitude limiter (hard limiter)  109  has its angle component extracted by frequency discriminator  115 . The output of frequency discriminator  115  has its DC component and random FM noise component removed by band-pass filter  116 . The output of band-pass filter  116  is integrated by integrator  117 , after which tangent function generator  118  generates the tangent value corresponding to the input angle.  
         [0041]    The output of delay circuit  114  is split into two portions, one of which is added by adder  120  to the output of constant generator  119 . The output of adder  120  is multiplied, by multiplier  121 , by the output of tangent function generator  118 . The output of multiplier  121  is split into two portions, one of which is supplied to band-pass filter  122  and the other to low-pass filter  123 . Band-pass filter  122  provides a signal from which unwanted noise components have been removed. This signal is supplied to matrix circuit  124 . The other split portion of the output of delay circuit  114  is also supplied to matrix circuit  124 , whereupon the left signal is output from left audio signal output terminal  125  and the right signal is output from right audio signal output terminal  126 . The pilot signal P is obtained from the output of low-pass filter  123 , and is output from pilot signal output terminal  127 .  
         [0042]    The operation of the component circuits will now be described using mathematical expressions. During its propagation, the signal radiated from transmitting antenna  101  is subject to random amplitude fluctuation and to phase fluctuation (termed“random FM noise”), which obey the Rayleigh distribution rule and can be represented respectively by ρ(t) and θ(t) in the amplitude and phase terms. These amplitude and phase fluctuations affect the signal as multiplicative disturbances. Hence the signal that arrives at C-Quam receiver antenna  102  is given by: 
           Srl ( t )=ρ( t )(1 +L+R ) cos (ω c   t +Φ( t )+θ( t ))  (4) 
         [0043]    The received signal is amplified by front-end amplifier  103  (which preferably changes its degree of amplification by means of a Received Signal Strength Indication mechanism). Then, by way of example, frequency converter  104  uses the signal from local oscillator  105 , which has a center angular frequency of ω c −ω 1  and an angular frequency fluctuation of ±δω, to convert the received signal to the difference frequency, whereby it is converted to an IF signal with a center angular frequency of al. IF filter  106  extracts just the required IF signal component. If thermal noise added by front-end amplifier  103  is ignored, the extracted signal is easily derived from Equation 4: 
           S 1 a ( t )=ρ( t )(1 +L+R ) cos ((ω 1 ±δω) t +Φ( t )+θ( t ))  (5) 
         [0044]    which can be rewritten as:  
               S1a        (   t   )       =       ρ        (   t   )            [       cos                   Θ        (   t   )         +       {         (       L   +     +     R   +       )          cos        (     Θ        (   t   )       )         -       H        (     (       L   +     +     R   +       )     )            sin        (     Θ        (   t   )       )           }     /   2     +       {         (       L   -     +     R   -       )          cos        (     Θ        (   t   )       )         +       H        (     (       L   -     +     R   -       )     )            sin        (     Θ        (   t   )       )           }     /   2       ]               (   6   )                               
 
         [0045]    where: 
         Θ( t )=(ω 1 ±δω) t +Φ( t )+θ( t ) 
         ( L   +   +R   + )=( L   −   +R   − ) 
           H ((L +   +R   + ))= H (( L   −   +R   − )) 
         [0046]    and (L + +R + ) and (L − +R − ) represent, respectively, the information signal present in the upper sideband region and the lower sideband region of the transmitted wave, and H((L + +R + )) represents the Hilbert transformation of (L + +R + ). The first, second and third terms in Equation 6 are the mathematical representations of the carrier component, the upper sideband component, and the lower sideband component, respectively. At this stage in the signal processing, the mutual arrangement of the upper and lower sideband components is the same as in the transmitted wave. From Equation 6 it will be seen that in an AM signal that is not overmodulated—i.e., in an AM signal that satisfies the condition expressed by Equation 3—the carrier component is always 6 dB higher than the sideband components. In FIG. 1, the signals are depicted in such manner that the upper sideband component and the lower sideband component can be distinguished. Although Equation 5 and Equation 6 are mathematically equivalent, when we are considering single sideband components we will use Equation 6 whenever it is necessary to discuss extracting a specific single sideband component, i.e., specifically either the upper sideband component or the lower sideband component.  
         [0047]    In this embodiment, we consider ordinary frequency conversion in a conventional AM receiver i.e., in an AM receiver for medium-frequency or high-frequency AM broadcasts) which includes a C-Quam receiver. Because a medium-frequency or high-frequency carrier is by its nature of relatively low frequency, it is often converted to an intermediate frequency ω IF1  using, as local oscillator frequency ω L1 , a frequency that is higher than the carrier frequency ω c . The purpose of this is to prevent admixture of spurious (unwanted) signals into the IF frequency region. If the sidebands of the received signal are observed when this is done, the upper and lower sidebands are seen to be reversed. If IF frequency ω IF1 , thus obtained is converted to a lower IF frequency cain and if this second frequency conversion is likewise performed using a frequency that is higher than IF frequency ω IF1 , the sidebands are again reversed and are thereby restored to their original arrangement. Although it is assumed that in practice this double conversion will be performed, in the present embodiment, for the sake of simplicity we have described a single-stage frequency conversion of the sort outlined in the preceding paragraphs, but this has no impact on the essence of the present invention. The same simplified version of frequency conversion is described in second and third embodiments of the invention below.  
         [0048]    The signal expressed by Equation 5 or Equation 6 is split into two portions. One portion of the split signal is supplied to frequency converter  107 , which uses local oscillator  108  with angular frequency ω 2  to form the sum frequency, thereby converting the signal from IF filter  106  to an IF signal with a center angular frequency of ω 1 +ω 2 . IF filter  110  then extracts only this required IF signal component. Using the representation given in Equation 6, this extracted signal can be given as:  
               S1b        (   t   )       =       ρ        (   t   )            [       cos        (       Θ        (   t   )       +       ω   2        t       )       +       {         (       L   +     +     R   +       )          cos        (       Θ        (   t   )       +       ω   2        t       )         -       H        (     (       L   +     +     R   +       )     )            sin        (       Θ        (   t   )       +       ω   2        t       )           }     /   2     +       {         (       L   -     +     R   -       )          cos        (       Θ        (   t   )       +       ω   2        t       )         +       H        (     (       L   -     +     R   -       )     )            sin        (       Θ        (   t   )       +       ω   2        t       )           }     /   2       ]               (   7   )                               
 
         [0049]    The other portion of the split signal is supplied to amplitude limiter (hard limiter)  109  where it is converted to a fixed-amplitude signal. Using the representation given in Equation 5, this is:  
               S1                 lim     =       cos        (         (       ω   1     ±   δω     )        t     +     Φ        (   t   )       +     θ        (   t   )         )       =     cos        (     Θ        (   t   )       )                 (   8   )                               
 
         [0050]    whereby it will be seen that the random amplitude fluctuation component ρ(t) has been removed. When the signals represented by Equation 7, i.e., the output of IF filter  110 , and by Equation 8, i.e., the output of amplitude limiter (hard limiter)  109 , are input to frequency converter  111  and their difference frequency component extracted, the signal obtained is:  
               S1c        (   t   )       =       ρ        (   t   )            [       cos        (       ω   2        t     )       +       {         (       L   +     +     R   +       )          cos        (       ω   2        t     )         -       H        (     (       L   +     +     R   +       )     )            sin        (       ω   2        t     )           }     /   2     +       {         (       L   -     +     R   -       )          cos        (       ω   2        t     )         +       H        (     (       L   -     +     R   -       )     )            sin        (       ω   2        t     )           }     /   2       ]               (   9   )               =       ρ        (   t   )            (     1   +   L   +   R     )          cos        (       ω   2        t     )                 (   10   )                               
 
         [0051]    In other words, the frequency fluctuation ±δω of local oscillator  105 , the modulation component Φ(t), and the random disturbance component θ(t) that are contained in the phase term can be completely removed. At the same time, the angular frequency of the carrier is converted to ω 2 . Consequently, frequency stability in the subsequent demodulation processing is dependent only on local oscillator  108 . As a result, if angular frequency ω 2  is low, frequency stability ceases to be a practical problem and a sharp filter can be used in the subsequent signal processing.  
         [0052]    Next, the use of IF filter  112  serves to extract, from the signal expressed by Equation 9, only the lower sideband signal, this being a signal from which some unwanted noise components have been removed and to which a carrier component has been added. Omitting remaining noise components from the mathematical expression, this extracted signal can be represented by: 
           S 1 d ( t )=ρ( t ){(1+( L   −   +R   − )/2) cos (ω 2   t )+( H (( L   − +R − )/2)) sin (ω 2   t )}  (11) 
         [0053]    which indicates that the lower sideband signal of the transmitted wave is extracted. Because this extracted lower sideband signal has a carrier component which, as mentioned previously, is 6 dB higher than the maximum value of the information signal, it can be used as ail RZ SSB signal. The use of RZ SSB demodulation processor  113  enables the random amplitude component ρ(t) to be removed and thereby provides a high-quality demodulated sum information signal (L+R).  
         [0054]    When the angle component is extracted by frequency discriminator  115  from the other split portion of the output of amplitude limiter (hard limiter)  109 , the resulting signal is:  
                     S1e        (   t   )       =          /          t        (     Θ        (   t   )       )                         =       (       ω   1     ±   δω     )     +          /          t        (       Φ        (   t   )       +     θ        (   t   )         )                             (   12   )                               
 
         [0055]    When the DC component and the random FM noise component contained in this signal are removed by band-pass filter  116 , the resulting signal is: 
           S 1 f ( t )= d/dt (Φ( t ))  (13) 
         [0056]    When this output of band-pass filter  116  is integrated by integrator  117 , the signal obtained is given by: 
           S 1 g ( t )=Φ( t )  (14) 
         [0057]    thereby providing an angle signal Φ(t) which contains the modulation component relating to the sum and difference signals. Tangent function generator  118  operates on this angle signal to form: 
           S 1 h ( t )=tan Φ( t )  (15) 
         [0058]    Delay circuit  114  is inserted to ensure that the processing delay up to and including RZ SSB demodulation processor  113  matches the processing delay from frequency discriminator  115  up to and including tangent function generator  118 . The output of delay circuit  114  is added by adder  120  to the output of constant generator  119 , whereby the following signal is obtained: 
           S 1 i ( t )=1 +L+R   (16) 
         [0059]    When the output of adder  120 , expressed by Equation 16, is multiplied in multiplier  121  by the output of tangent function generator  118 , expressed by Equation 15, the difference signal (L−R+P) is obtained:  
                     S1j        (   t   )       =       (     1   +   L   +   R     )        tan                   Φ        (   t   )                     =     L   -   R   +   P                   (   17   )                               
 
         [0060]    The relation shown in Equation 2 is used to derive this. The output of multiplier  121  is split into two portions, one of which is supplied to band-pass filter  122  and the other to low-pass filter  123 . The difference signal (L−R) from which unwanted noise components have been removed by band-pass filter  122 , and the sum signal (L+R) obtained from delay circuit  114 , are supplied to matrix circuit  124 , whereupon the left signal is output from left audio signal output terminal  125  and the right signal is output from right audio signal output terminal  126 . The pilot signal P is obtained from the output of low-pass filter  123 , and is output from pilot signal output terminal  127 .  
         [0061]    The signal processing after IF filter  106  can be performed by a DSP circuit. As explained above, when the lower sideband signal with added carrier component is extracted, frequency stability is determined solely by local oscillator  108 ,  7508  and therefore this extraction can be performed using IF filter  112  having sharp cut-off characteristics. Other advantages of a filter implemented by a DSP circuit include the fact that temperature characteristics, etc., do not have to be taken into consideration. If the embodiment shown in FIG. 1 is implemented using a DSP device, in order to make the frequency region in which unnecessary processing is performed as small as possible and so reduce the DSP power consumption, it is necessary to lower the sampling frequency of the RZ SSB demodulation processor. This can be achieved by shifting the signal frequency region to as low a frequency region as possible.  
         [0062]    Second Embodiment  
         [0063]    A second specific embodiment of the present invention will now be described. FIG. 2 is a block diagram showing the configuration of this second embodiment, which comprises C-Quam transmitter  200 , transmitting antenna  201 , receiving antenna  202  of a C-Quam receiver, front-end amplifier  203 , frequency converter  204 , local oscillator  205 , IF filter  206 , frequency converter  207 , local oscillator  208 , amplitude limiter (hard limiter)  209 , IF filter  210 , frequency converter  211 , IF filter  212 , RZ SSB demodulation processor  213 , delay circuit  214 , frequency discriminator  215 , band-pass filter  216 , integrator  217 , tangent function generator  218 , constant generator  219 , adder  220 , multiplier  221 , band-pass filter  222 , low-pass filter  223 , matrix circuit  224 , left audio signal output terminal  225 , right audio signal output terminal  226 , and pilot signal output terminal  227 .  
         [0064]    A brief description will now be given of signal flow in this second embodiment shown in FIG. 2, and of the functioning of its component circuits.  
         [0065]    The output of C-Quam transmitter  200  is transmitted as a C-Quam modulated wave by transmitting antenna  201 .  
         [0066]    The C-Quam modulated wave is received by antenna  202  of the C-Quam receiver, and after it has been amplified by front-end amplifier  203 , is converted by frequency converter  204  and local oscillator  205  to the difference frequency signal, whereupon the required IF signal is extracted by IF filter  206 .  
         [0067]    The extracted signal is split into two portions and one portion supplied to frequency converter  207 . The difference frequency between this signal and the signal from local oscillator  208  is extracted by IF filter  210 . The other split portion of the signal is supplied to amplitude limiter (hard limiter)  209  where it is converted to a fixed-amplitude signal. The output of amplitude limiter (hard limiter)  209  is split into two portions and one portion supplied to frequency converter  211 , which functions so as to form the sum frequency component from the output of frequency limiter  209  and the output of IF filter  210 . IF filter  212  extracts the lower sideband component from the output signal of frequency converter  211 , this lower sideband component having had some unwanted noise components removed and being accompanied by a carrier component. The output of IF filter  212  is supplied to RZ SSB demodulation processor  213  which demodulates it, thereby obtaining the sum signal (L+R). This sum signal is supplied to delay circuit  214 .  
         [0068]    The other split portion of the output of amplitude limiter (hard limiter)  209  has its angle component extracted by frequency discriminator  215 . The output of frequency discriminator  215  has its DC component and random FM noise component removed by band-pass filter  216 . The output of band-pass filter  216  is integrated by integrator  217 , after which tangent function generator  218  generates the tangent value corresponding to the input angle.  
         [0069]    The output of delay circuit  214  is split into two portions, one of which is added by adder  220  to the output of constant generator  219 . The output of adder  220  is multiplied, by multiplier  221 , by the output of tangent function generator  218 . The output of multiplier  221  is split into two portions, one of which is supplied to band-pass filter  222  and the other to low-pass filter  223 . Band-pass filter  222  provides a signal from which unwanted noise components have been removed. This signal is supplied to matrix circuit  224 . The other split portion of the output of delay circuit  214  is also supplied to matrix circuit  224 , whereupon the left signal is output from left audio signal output terminal  225  and the right signal is output from right audio signal output terminal  226 . The pilot signal P is obtained from the output of low-pass filter  223 , and is output from pilot signal output terminal  227 .  
         [0070]    The operation of the component circuits will now be described using mathematical expressions. During its propagation, the signal radiated from transmitting antenna  201  is subject to random amplitude fluctuation and to phase fluctuation (random FM noise) which obey the Rayleigh distribution rule and can be represented respectively by ρ(t) and θ(t) in the amplitude and phase terms. These amplitude and phase fluctuations affect the signal as multiplicative disturbances. Hence the signal that arrives at C-Quam receiver antenna  202  is given by: 
           Sr 2( t )=ρ( t )(1 +L+R ) cos (ω c   t +Φ( t )+θ( t ))  (18) 
         [0071]    The received signal is amplified by front-end amplifier  203  (which preferably changes its degree of amplification by means of a Received Signal Strength Indication mechanism). Then, by way of example, frequency converter  204  uses the signal from local oscillator  205 , which has a center angular frequency of ω c −ω 1  and an angular frequency fluctuation of ±δω, to convert the received signal to the difference frequency, whereby it is converted to an IF signal with a center angular frequency of ω 1 . IF filter  206  extracts just the required IF signal component. If thermal noise added by front-end amplifier  203  is ignored, the extracted signal is easily derived from Equation 18: 
           S 2 a ( t )=ρ( t )(1 +L+R ) cos ((ω 1 ±δ) t +Φ( t )+θ( t ))  (19) 
         [0072]    which can be rewritten as:  
               S2a        (   t   )       =       ρ        (   t   )            [       cos                   Θ        (   t   )         +       {         (       L   +     +     R   +       )          cos        (     Θ        (   t   )       )         -       H        (     (       L   +     +     R   +       )     )            sin        (     Θ        (   t   )       )           }     /   2     +       {         (       L   -     +     R   -       )          cos        (     Θ        (   t   )       )         +       H        (     (       L   -     +     R   -       )     )            sin        (     Θ        (   t   )       )           }     /   2       ]               (   20   )                               
 
         [0073]    where: 
         Θ( t )=(ω 1 ±δω) t +Φ( t )+θ( t ) 
         ( L   +   +R   + )=( L   −   +R   − ) 
           H (( L   +   +R   + ))= H (( L   −   +R   − )) 
         [0074]    and (L + +R + ) and (L − +R − ) represent, respectively, the information signal present in the upper sideband region and the lower sideband region of the transmitted wave, and H((L + +R + )) represents the Hilbert transformation of (L + +R + ). The first, second and third terms in Equation 20 are the mathematical representations of the carrier component, the upper sideband component, and the lower sideband component, respectively. At this stage min the signal processing, the mutual arrangement of the upper and lower sideband components is the same as in the transmitted wave. From Equation 20 it will be seen that in an AM signal that is not overmodulated—i.e., in an AM signal that satisfies the condition expressed by Equation 3—the carrier component is always 6 dB higher than the sideband components. In FIG. 2, the signals are depicted in such manner that the upper sideband component and the lower sideband component can be distinguished. Although Equation 19 and Equation 20 are mathematically equivalent, when we are considering single sideband components we will use Equation 20 whenever it is necessary to discuss extracting a specific single sideband component, i.e., specifically either the upper sideband component or the lower sideband component.  
         [0075]    The signal expressed by Equation 19 or Equation 20 is split into two portions. One portion of the split signal is supplied to frequency converter  207 , which uses local oscillator  208  with angular frequency ω 2  to form the difference frequency, thereby converting the signal from IF filter  206  to an IF signal with a center angular frequency of ω 2 −ω 1 . IF filter  210  then extracts only this required IF signal component. This extracted signal can be given as:  
                     S2b        (   t   )       =                  ρ        (   t   )            (     1   +   L   +   R     )          cos        (         ω   2        t     -     Θ        (   t   )         )                     =                  ρ        (   t   )       [       cos        (         ω   2        t     -     Θ        (   t   )         )       +     {         (       L   +     +     R   +       )          cos        (         ω   2        t     -     Θ        (   t   )         )         +                                            H        (     (       L   +     +     R   +       )     )            sin        (         ω   2        t     -     Θ        (   t   )         )         }       /   2     +     {     (       L   -     +     R   -       )                                        cos        (         ω   2        t     -     Θ        (   t   )         )       -       H        (     (       L   -     +     R   -       )     )            sin        (         ω   2        t     -     Θ        (   t   )         )           }       /   2     ]                 (   21   )                               
 
         [0076]    In this embodiment, the frequency relation ω 2 &gt;ω 1  is used. It will be seen that the positions of the top and bottom sideband components of the transmitted wave are reversed.  
         [0077]    The other portion of the split signal is supplied to amplitude limiter (hard limiter)  209  where it is converted to a fixed-amplitude signal. Using the representation given in Equation 19, this is:  
                     S2                 lim     =     cos        (         (       ω   1     ±   δω     )        t     +     Φ        (   t   )       +     θ        (   t   )         )                   =     cos        (     Θ        (   t   )       )                     (   22   )                               
 
         [0078]    whereby it will be seen that the random amplitude fluctuation component ρ(t) has been removed. When the signals represented by Equation 21, i.e., the output of IF filter  210 , and by Equation 22, i.e., the output of amplitude limiter (hard limiter)  209 , are input to frequency converter  211  and their sum frequency component extracted, the signal obtained is:  
               S2c        (   t   )       =       ρ        (   t   )            [       cos        (       ω   2        t     )       +       {         (       L   +     +     R   +       )          cos        (       ω   2        t     )         +       H        (     (       L   +     +     R   +       )     )            sin        (       ω   2        t     )           }     /   2     +       {         (       L   -     +     R   -       )          cos        (       ω   2        t     )         -       H        (     (       L   -     +     R   -       )     )            sin        (       ω   2        t     )           }     /   2       ]               (   23   )               =       ρ        (   t   )            (     1   +   L   +   R     )          cos        (       ω   2        t     )                 (   24   )                               
 
         [0079]    In other words, the frequency fluctuation ±δω of local oscillator  205 , the modulation component Φ(t), and the random disturbance component θ(t) that were present in the phase term can be completely removed. At the same time, the angular frequency of the carrier is converted to ω 2 . Consequently, frequency stability in the subsequent demodulation processing is dependent only on local oscillator  208 . As a result, if angular frequency ω 2  is low, frequency stability ceases to be a practical problem and a sharp filter can be used in the subsequent signal processing.  
         [0080]    The use of IF filter  212  serves to extract, from the signal output by frequency converter  211  and expressed by Equation 23—this being a signal from which some unwanted noise components have been removed and to which a carrier component has been added—only the lower sideband signal. Omitting remaining noise components from the mathematical expression, this extracted signal can be derived from Equation 23 and represented by: 
           S 2 d ( t )=ρ( t ){(1+( L   +   +R   + )/2) cos (ω 2   t )+( H (( L   +   +R   + )/2)) sin (ω 2   t )}  (25) 
         [0081]    As previously mentioned, the extracted lower sideband signal that can be described by Equation 25 corresponds to the upper sideband of the transmitted wave. Because this extracted lower sideband signal has a carrier component which is 6 dB higher than the maximum value of the information signal, it can be used as an RZ SSB signal. The use of RZ SSB demodulation processor  213  enables the random amplitude component ρ(t) to be removed and thereby provides a high-quality demodulated sum information signal (L+R).  
         [0082]    When the angle component is extracted by frequency discriminator  215  from the other split portion of the output of amplitude limiter (hard limiter)  209 , the resulting signal is:  
                     S2e        (   t   )       =          /          t        (     Θ        (   t   )       )                         =       (       ω   1     ±   δω     )     +          /          t        (       Φ        (   t   )       +     θ        (   t   )         )                             (   26   )                               
 
         [0083]    When the DC component and the random FM noise component contained in this signal are removed by band-pass filter  216 , the resulting signal is: 
           S 2 f ( t )= d/dt (Φ( t ))  (27) 
         [0084]    When this output of band-pass filter  216  is integrated by integrator  217 , the signal obtained is given by: 
           S 2 g ( t )=Φ( t )  (28) 
         [0085]    thereby providing an angle signal Φ(t) which contains the modulation component relating to the sum and difference signals. Tangent function generator  218  operates on this angle signal to form: 
           S 2 h ( t )=tan Φ( t )  (29) 
         [0086]    Delay circuit  214  is inserted to ensure that the processing delay up to and including RZ SSB demodulation processor  213  matches the processing delay from frequency discriminator  215  up to and including tangent function generator  218 . The output of delay circuit  214  is added by adder  220  to the output of constant generator  219 , whereby the following signal is obtained: 
           S 2 i ( t )=1 +L+R   (30) 
         [0087]    When the output of adder  220 , expressed by Equation 30, is multiplied in multiplier  221  by the output of tangent function generator  218 , expressed by Equation 29, the difference signal (L−R+P) is obtained:  
                     S2j        (   t   )       =       (     1   +   L   +   R     )        tan                   Φ        (   t   )                     =     L   -   R   +   P                   (   31   )                               
 
         [0088]    The relation shown in Equation 2 is used to derive this. The output of multiplier  221  is split into two portions, one of which is supplied to band-pass filter  222  and the other to low-pass filter  223 . The difference signal (L−R) from which unwanted noise components have been removed by band-pass filter  222 , and the sum signal (L+R) obtained from delay circuit  214 , are supplied to matrix circuit  224 , whereupon the left signal is output from left audio signal output terminal  225  and the right signal is output from right audio signal output terminal  226 . The pilot signal P is obtained from the output of low-pass filter  223 , and is output from pilot signal output terminal  227 .  
         [0089]    The signal processing after IF filter  206  can be performed by a DSP circuit. As explained above, when the lower sideband signal with added carrier component is extracted, frequency stability is determined solely by local oscillator  208 , and therefore this extraction can be performed using IF filter  212  having sharp cut-off characteristics. Other advantages of a filter implemented by a DSP circuit include the fact that temperature characteristics, etc., do not have to be taken into consideration. If the embodiment shown in FIG. 2 is implemented using a DSP device, in order to make the frequency region in which unnecessary processing is performed as small as possible and so reduce the DSP power consumption, it is necessary to lower the sampling frequency of the RZ SSB demodulation processor. This can be achieved by shifting the signal frequency region to as low a frequency region as possible.  
         [0090]    Third Embodiment  
         [0091]    A third specific embodiment of the present invention will now be described. FIG. 3 is a block diagram showing the configuration of this third embodiment, which comprises C-Quam transmitter  300 , transmitting antenna  301 , receiving antenna  302  of a C-Quam receiver, front-end amplifier  303 , frequency converter  304 , local oscillator  305 , IF filter  306 , frequency converter  307 , local oscillator  308 , amplitude limiter (hard limiter)  309 , IF filters  310  and  311 , frequency converters  312  and  313 , adder  314 , IF filter  315 , RZ SSB demodulation processor  316 , delay circuit  317 , frequency discriminator  318 , band-pass filter  319 , integrator  320 , tangent function generator  321 , constant generator  322 , adder  323 , multiplier  324 , band-pass filter  325 , low-pass filter  326 , matrix circuit  327 , left audio signal output terminal  328 , right audio signal output terminal  329 , and pilot signal output terminal  330 .  
         [0092]    A brief description will now be given of signal flow in this third embodiment shown in FIG. 3, and of the functioning of its component circuits.  
         [0093]    The output of C-Quam transmitter  300  is transmitted, as a C-Quam modulated wave by transmitting antenna  301 .  
         [0094]    The C-Quam modulated wave is received by antenna  302  of the C-Quam receiver, and after it has been amplified by front-end amplifier  303 , is converted by frequency converter  304  and local oscillator  305  to the difference frequency signal, whereupon the required IF signal is extracted, by IF filter  306 .  
         [0095]    The extracted signal is split into two portions and one portion supplied to frequency converter  307 . The sum and difference frequencies between this signal and the signal from local oscillator  308  are formed. The sum frequency is extracted by IF filter  310  and the difference frequency is extracted by IF filter  311 . The other split portion of the signal is supplied to amplitude limiter (hard limiter)  309  where it is converted to a fixed-amplitude signal. The output of amplitude limiter (hard limiter)  309  is split into two portions, and one portion is again split into two portions. Frequency converter  312  uses one of the split output portions from amplitude limiter  309  to form the difference frequency component between this signal and the output signal of IF filter  310 . Likewise, frequency converter  313  uses one of the split output portions from amplitude limiter  309  to form the sum frequency component between this signal and the output signal of IF filter  311 . The outputs of frequency converter  312  and frequency converter  313  are added by adder  314 , whereupon IF filter  315  extracts the lower sideband component, this being a signal from which some unwanted noise components have been removed and which is accompanied by a carrier component. The output of IF filter  315  is supplied to RZ SSB demodulation processor  316  which demodulates it, thereby obtaining the sum signal (L+R). This sum signal is supplied to delay circuit  317 .  
         [0096]    The other split portion of the output of amplitude limiter (hard limiter)  309  has its angle component extracted by frequency discriminator  318 . The output of frequency discriminator  318  has its DC component and random FM noise component removed by band-pass filter  319 . The output of band-pass filter  319  is integrated by integrator  320 , after which tangent function generator  321  generates the tangent value corresponding to the input angle.  
         [0097]    The output of delay circuit  317  is split into two portions, one of which is added by adder  323  to the output of constant generator  322 . The output of adder  323  is multiplied, by multiplier  324 , by the output of tangent function generator  321 . The output of multiplier  324  is split into two portions, one of which is supplied to band-pass filter  325  and the other to low-pass filter  326 . Band-pass filter  325  provides a signal from which unwanted noise components have been removed. This signal is supplied to matrix circuit  327 . The other split portion of the output of delay circuit  317  is also supplied to matrix circuit  327 , whereupon the left signal is output from left audio signal output terminal  328  and the right signal is output from right audio signal output terminal  329 . The pilot signal P is obtained from the output of low-pass filter  326 , and is output from pilot signal output terminal  330 .  
         [0098]    The operation of the component circuits will now be described using mathematical expressions. During its propagation, the signal radiated from transmitting antenna  301  is subject to random amplitude fluctuation and to phase fluctuation (random FM noise) which obey the Rayleigh distribution rule and can be represented respectively by ρ(t) and θ(t) in the amplitude and phase terms. These amplitude and phase fluctuations affect the signal as multiplicative disturbances. Hence the signal that arrives at C-Quam receiver antenna  302  is given by: 
           Sr 3( t )=ρ( t )(1 +L+R ) cos (ω c   t+Φ ( t )+θ( t ))  (32) 
         [0099]    The received signal is amplified by front-end amplifier  303  (which preferably changes its degree of amplification by means of a Received Signal Strength Indication mechanism). Then, by way of example, frequency converter  304  uses the signal from local oscillator  305 , which has a center angular frequency of ω c −ω 1  and an angular frequency fluctuation of ±δω, to convert the received signal to the difference frequency, whereby it is converted to an IF signal with a center angular frequency of ω 1 . IF filter  306  extracts just the required IF signal component. If thermal noise added by front-end amplifier  303  is ignored, the extracted signal is easily derived from Equation 32: 
           S 3 a ( t )=ρ( t )(1 +L+R ) cos ((ω 1 ±δω) t +Φ( t )+θ( t ))  (33) 
         [0100]    which can be rewritten as:  
               S3a        (   t   )       =       ρ        (   t   )            [       cos                   Θ        (   t   )         +       {         (       L   +     +     R   +       )          cos        (     Θ        (   t   )       )         -       H        (     (       L   +     +     R   +       )     )            sin        (     Θ        (   t   )       )           }     /   2     +       {         (       L   -     +     R   -       )          cos        (     Θ        (   t   )       )         +       H        (     (       L   -     +     R   -       )     )            sin        (     Θ        (   t   )       )           }     /   2       ]               (   34   )                               
 
         [0101]    where: 
         Θ( t )=(ω 1 ±δω) t +Φ( t )+θ( t ) 
         ( L   +   +R   + )=( L   −   +R   − ) 
           H (( L   +   +R   + ))= H (( L   −   +R   − )) 
         [0102]    and (L + +R + ) and (L − +R − ) represent, respectively, the information signal present in the upper sideband region and in the lower sideband region of the transmitted wave, and H((L + +R + )) represents the Hilbert transformation of (L + +R + ). The first, second and third terms in Equation 34 are the mathematical representations of the carrier component, the upper sideband component, and the lower sideband component, respectively. At this stage in the signal processing, the mutual arrangement of the upper and lower sideband components is the same as in the transmitted wave. From Equation 34 it will be seen that in an AM signal that is not overmodulated—i.e., in an AM signal that satisfies the condition expressed by Equation 3—the carrier component is always 6 dB higher than the sideband components. In FIG. 3, the signals are depicted in such manner that the upper sideband component and the lower sideband component can be distinguished. Although Equation 33 and Equation 34 are mathematically equivalent, when we are considering single sideband components we will use Equation 34 whenever it is necessary to discuss extracting a specific single sideband component, i.e., specifically either the upper sideband component or the lower sideband component.  
         [0103]    The signal expressed by Equation 33 or Equation 34, and which is the output of IF filter  306 , is split into two portions. One portion of the split signal is supplied to frequency converter  307 , which uses local oscillator  308  with angular frequency ω 2  to form the sum frequency, thereby converting the signal from IF filter  306  to an IF signal with a center angular frequency of ω 1 +ω 2 . IF filter  310  then extracts only this required IF signal component. Using Equation 34, this extracted signal can be given as:  
               S3b        (   t   )       =       ρ        (   t   )            [       cos        (       Θ        (   t   )       +       ω   2        t       )       +       {         (       L   +     +     R   +       )          cos        (       Θ        (   t   )       +       ω   2        t       )         -       H        (     (       L   +     +     R   +       )     )            sin        (       Θ        (   t   )       +       ω   2        t       )           }     /   2     +       {         (       L   -     +     R   -       )          cos        (       Θ        (   t   )       +       ω   2        t       )         +       H        (     (       L   -     +     R   -       )     )            sin        (       Θ        (   t   )       +       ω   2        t       )           }     /   2       ]               (   35   )                               
 
         [0104]    Frequency converter  307  also uses local oscillator  308  with angular frequency ω 2  to form the difference frequency, thereby converting the signal from IF filter  306  to an IF signal with a center angular frequency of ω 2 −ω 1 . IF filter  311  then extracts only this required IF signal component. This signal is:  
                     S3c        (   t   )       =       ρ        (   t   )       [       cos        (         ω   2        t     -     Θ        (   t   )         )       +       (       L   +     +     R   +       )          cos        (         ω   2        t     -     Θ        (   t   )         )         -       H        (     (       L   +     +     R   +       )     )            sin        (         ω   2        t     -     Θ        (   t   )         )               }     /   2     +       {         (       L   -     +     R   -       )          cos        (         ω   2        t     -     Θ        (   t   )         )         +       H        (     (       L   -     +     R   -       )     )            sin        (         ω   2        t     -     Θ        (   t   )         )           }     /   2             (   36   )                               
 
         [0105]    In this embodiment, the frequency relation ω 2 &gt;ω 1  is used, and hence the positions of the top and bottom sideband components of the transmitted wave are reversed in the output of IF filter  311 .  
         [0106]    The other portion of the split signal from IF filter  306  is supplied to amplitude limiter (hard limiter)  309  where it is converted to a fixed-amplitude signal, given by:  
                     S3                 lim     =     cos        (         (       ω   1     ±   δω     )        t     +     Φ        (   t   )       +     θ        (   t   )         )                   =     cos        (     Θ        (   t   )       )                     (   37   )                               
 
         [0107]    whereby it will be seen that the random amplitude fluctuation component ρ(t) has been removed.  
         [0108]    When the signals represented by Equation 35, i.e., the output of IF filter  310 , and by Equation 37, i.e., the output of amplitude limiter (hard limiter)  309 , are input to frequency converter  312  and their difference frequency component extracted, the signal obtained is:  
               S3d        (   t   )       =       ρ        (   t   )            [       cos        (       ω   2        t     )       +       {         (       L   +     +     R   +       )          cos        (       ω   2        t     )         -       H        (     (       L   +     +     R   +       )     )            sin        (       ω   2        t     )           }     /   2     +       {         (       L   -     +     R   -       )          cos        (       ω   2        t     )         +       H        (     (       L   -     +     R   -       )     )            sin        (       ω   2        t     )           }     /   2       ]               (   38   )                               
 
         [0109]    Likewise, when the signals represented by Equation 36, i.e., the output of IF filter  311 , and by Equation 37, i.e., the output of amplitude limiter (hard limiter)  309 , are input to frequency converter  313  and their sum frequency component extracted, the signal obtained is:  
               S3e        (   t   )       =       ρ        (   t   )            [       cos        (       ω   2        t     )       +       {         (       L   +     +     R   +       )          cos        (       ω   2        t     )         +       H        (     (       L   +     +     R   +       )     )            sin        (       ω   2        t     )           }     /   2     +       {         (       L   -     +     R   -       )          cos        (       ω   2        t     )         -       H        (     (       L   -     +     R   -       )     )            sin        (       ω   2        t     )           }     /   2       ]               (   39   )                               
 
         [0110]    Equations 38 and 39 indicate that the frequency fluctuation ±δω of local oscillator  305 , the modulation component Φ(t), and the random disturbance component θ(t) that were present in the phase term can be completely removed. At the same time, the angular frequency of the carrier is converted to ω 2 . Consequently, frequency stability in the subsequent demodulation processing is dependent only on local oscillator  308 . As a result, if angular frequency ω 2  is low, frequency stability ceases to be a practical problem and a sharp filter can be used in the subsequent signal processing.  
         [0111]    The outputs of frequency converters  312  and  313 , which can be described by Equations 38 and 39, are added by adder  314 , whereupon use of IF filter  315  serves to extract only the lower sideband signal, which is a signal from which some unwanted noise components have been removed and to which a carrier component has been added. Omitting remaining noise components from the mathematical expression, this extracted signal can be represented by:  
               S3f        (   t   )       =       ρ        (   t   )            [       2        cos        (       ω   2        t     )         +       {         (       L   +     +     R   +       )          cos        (       ω   2        t     )         +       H        (     (       L   +     +     R   +       )     )            sin        (       ω   2        t     )           }     /   2     +       {         (       L   -     +     R   -       )          cos        (       ω   2        t     )         +       H        (     (       L   -     +     R   -       )     )            sin        (       ω   2        t     )           }     /   2       ]               (   40   )                               
 
         [0112]    Because the second and third terms of Equation 40 were originally the upper and lower sidebands, respectively, when the transmitted wave propagated through the propagation path, a diversity effect can be anticipated, since the upper and lower sidebands can be expected to experience different degrees of deterioration during propagation. As mentioned in the description of the first embodiment, it is evident that the lower sideband signal given by Equation 40 can be used as an RZ SSB signal. Accordingly, the use of an RZ SSB demodulation processor enables the disturbance component ρ(t) to be removed, which, coupled with the diversity effect, enables a high-quality demodulated sum signal (L+R) to be obtained.  
         [0113]    When the angle component is extracted by frequency discriminator  318  from the other split portion of the output of amplitude limiter (hard limiter)  309 , the resulting signal is:  
                     S3g        (   t   )       =          /          t        (     Θ        (   t   )       )                         =       (       ω   1     ±   δω     )     +          /          t        (       Φ        (   t   )       +     θ        (   t   )         )                             (   41   )                               
 
         [0114]    When the DC component and the random FM noise component contained in this signal are removed by band-pass filter  319 , the resulting signal is: 
           S 3 h ( t )= d/dt (Φ( t ))  (42) 
         [0115]    When this output of band-pass filter  319  is integrated by integrator  320 , the signal obtained is given by: 
           S 3 i ( t )=Φ( t )  (43) 
         [0116]    thereby providing an angle signal Φ(t) which contains the modulation component relating to the sum and difference signals. Tangent function generator  321  operates on this angle signal to form: 
           S 3 j ( t )=tan Φ( t )  (44) 
         [0117]    Delay circuit  317  is inserted to ensure that the processing delay up to and including RZ SSB demodulation processor  316  matches the processing delay from frequency discriminator  318  up to and including tangent function generator  321 . The output of delay circuit  317  is added by adder  323  to the output of constant generator  322 , whereby the following signal is obtained: 
           S 3 k ( t )=1 +L+R   (45) 
         [0118]    When the output of adder  323 , expressed by Equation 45, is multiplied in multiplier  324  by the output of tangent function generator  321 , expressed by Equation 44, the difference signal (L−R+P) is obtained:  
                     S3l        (   t   )       =       (     1   +   L   +   R     )        tan                   Φ        (   t   )                     =     L   -   R   +   P                   (   46   )                               
 
         [0119]    The relation shown in Equation 2 is used to derive this. The output of multiplier  324  is split into two portions, one of which is supplied to band-pass filter  325  and the other to low-pass filter  326 . The difference signal (L−R) from which unwanted noise components have been removed by band-pass filter  325 , and the sum signal (L+R) obtained from delay circuit  317 , are supplied to matrix circuit  327 , whereupon the left signal is output from left audio signal output terminal  328  and the right signal is output from right audio signal output terminal  329 . The pilot signal P is obtained from the output of low-pass filter  326 , and is output from pilot signal output terminal  330 .  
         [0120]    The signal processing after IF filter  306  can be performed by a DSP circuit. As explained above, when the lower sideband signal with added carrier component is extracted, frequency stability is determined solely by local oscillator  308 , and therefore this extraction can be performed using IF filter  315  having sharp cut-off characteristics. Other advantages of a filter implemented by a DSP circuit include the fact that temperature characteristics, etc., do not have to be taken into consideration. If the embodiment shown in FIG. 3 is implemented using a DSP device, in order to make the frequency region in which unnecessary processing is performed as small as possible and so reduce the DSP power consumption, it is necessary to lower the sampling frequency of the RZ SSB demodulation processor. This can be achieved by shifting the signal frequency region to as low a frequency region as possible.  
         [0121]    As has been described above, the present invention provides the following benefits: 
         [0122]    1. A demodulated signal with frequency characteristics that are faithful to the frequency characteristics of the transmitted wave is obtained, and the quality of the demodulated signal is better than that obtained with a conventional receiving circuit.  
         [0123]    2. Reception characteristics are resistant to external multiplicative noise resulting from fading and so forth, and hence the quality of the demodulated signal is improved.  
         [0124]    3. The invention provides a receiving circuit configuration which, while maintaining the advantages of a conventional AM receiver, can demodulate a C-Quam modulated signal without being strongly affected by frequency fluctuations. The receiver of the invention therefore has an inexpensive configuration.  
         [0125]    4. An improvement in demodulation quality is achieved by configuring the receiving circuit so that, by using both the upper sideband and the lower sideband obtained by processing a C-Quam modulated signal, a frequency diversity effect is obtained.