Abstract:
A method and circuit for timing the start of a precharge period in an eDRAM. The circuit including: a delayed lock loop circuit for receiving a clock signal and generating a control signal for adjusting an internal delay of the clock signal; and means for generating a delayed clock signal in response to the control signal. The means for generating the delayed clock signal is a multiple stage delay circuit, each stage of the multiple delay stage circuit connected in series and each stage individually responsive to the control signal.

Description:
BACKGROUND OF INVENTION 
   1. Field of the Invention 
   The present invention relates to the field of integrated circuits; more specifically, it relates to a circuit and a method for precise timing of signals in an embedded dynamic access memory array (eDRAM). 
   2. Background of the Invention 
   As eDRAM operational frequencies continue to increase, the duration of the precharge period is scaling faster than the duration of the active period of the read/write cycle making it more difficult to time the start of the precharge period. This can result in insufficient signal development for subsequent read cycles and/or a decrease in data retention. 
   SUMMARY OF INVENTION 
   A first aspect of the present invention is a circuit for timing the start of a precharge period in an eDRAM comprising: a delayed lock loop circuit for receiving a clock signal and generating a control signal for adjusting an internal delay of the clock signal; and means for generating a delayed clock signal in response to the control signal. 
   A second aspect of the present invention is a method for timing the start of a precharge period in an eDRAM comprising: providing a delayed lock loop circuit for receiving a clock signal and generating a control signal for adjusting an internal delay of the clock signal; and providing means for generating a delayed clock signal in response to the control signal. 
   A third aspect of the present invention is an eDRAM comprising: an array of memory cells interconnected by wordlines and bitlines; a delayed lock loop circuit for receiving a clock signal and generating a control signal for adjusting an internal delay of the clock signal; means for generating first, second, third and fourth delayed clock signals in response to the control signal; a wordline driver for activating wordlines in the eDRAM in response to an address signal, the wordline driver responsive to the third delayed clock signal; a sense amplifier circuit for amplifying data signals on the bitlines, the sense amplifier circuit responsive to the first delayed clock signal and the second delayed clock signal; a bitline precharge circuit for precharging the bitlines, the bitline recharge circuit responsive to the third delayed clock; and a column select circuit for selecting particular bitlines to connect to means for outputting data signals from the array, the column select circuit responsive to the second delayed clock signal and the means for outputting data signals responsive to the fourth delayed clock signal. 
   A fourth aspect of the present invention is a method of synchronous control of an eDRAM comprising an array of memory cells interconnected by wordlines and bitlines, the method comprising: providing a delayed lock loop circuit for receiving a clock signal and generating a control signal for adjusting an internal delay of the clock signal; providing means for generating first, second, third and fourth delayed clock signals in response to the control signal; providing a wordline driver for activating wordlines in the eDRAM in response to an address signal, the wordline driver responsive to the third delayed clock signal; providing a sense amplifier circuit for amplifying data signals on the bitlines, the sense amplifier circuit responsive to the first delayed clock signal and the second delayed clock signal; providing a bitline precharge circuit for precharging the bitlines, the bitline recharge circuit responsive to the third delayed clock; and providing a column select circuit for selecting particular bitlines to connect to means for outputting data signals from the array, the column select circuit responsive to the second delayed clock signal and the means for outputting data signals responsive to the fourth delayed clock signal. 

   
     BRIEF DESCRIPTION OF DRAWINGS 
     The features of the invention are set forth in the appended claims. The invention itself, however, will be best understood by reference to the following detailed description of an illustrative embodiment when read in conjunction with the accompanying drawings, wherein: 
       FIG. 1  is an exemplary schematic circuit diagram of an eDRAM array; 
       FIG. 2  is a timing diagram for the e-eDRAM of  FIG. 1 ; 
       FIG. 3  is a block schematic diagram of a circuit for precise timing control of the start of the restore period of the eDRAM of  FIG. 1  according to a first embodiment of the present invention; 
       FIG. 4  is a schematic diagram of an exemplary delay element of the circuit of  FIG.3 ; 
       FIG. 5  is a schematic diagram of an exemplary fine-tuning circuit of the circuit of  FIG. 3 ; 
       FIG. 6  is a schematic diagram of an exemplary fuse bank for generating control signals for the circuit of  FIG. 3 ; 
       FIG. 7  is a block schematic diagram of an eDRAM according to a second embodiment of the present invention; and 
       FIG. 8  is a block schematic diagram of an eDRAM according to a third embodiment of the present invention. 
   

   DETAILED DESCRIPTION 
   The term and symbol V DD  indicates the logic high voltage and logic and array signals swing between V DD  and ground unless otherwise noted. The term and symbol V REFX  indicates reference cell precharge voltage, which, in one example, has a value equal to the value of V DD /2. The term storage capacitor may be read as storage node, since the present invention is not necessarily limited to capacitive storage devices. The terms precharge and restore are equivalent terms and the terms timing and clocking are equivalent terms. A timing phase or a clock phase refers to a portion of a whole clock cycle from a high to the next high or from a low to the next low being 360°. For example, a phase of 90° is a quarter of a clock cycle, a phase of 180° is half a clock cycle and a phase of 270° is three-quarters of a clock cycle. A delay in a synchronous signal is a phase shift of that signal and the terms delay and phase shift may be used interchangeably. Delay may be in units of time or degrees of phase. 
     FIG. 1  is an exemplary schematic circuit diagram of an eDRAM array  100 . In  FIG. 1 , eDRAM array  100  includes a data cell array  105 , a reference cell array  110 , a sense amplifier section  115  and column select (CS) circuits  120 . eDRAM array  100  is a GND restore eDRAM, i.e. the bitlines (BLs) are initialized to GND. For simplicity, eDRAM  100  includes only four wordlines WL 0 , WL 1 , WL 2  and WL 3  and two bitline pairs BT 0  and BC 0  (true and complement of bitline  0 ) and BT 1  and BC 1  (true and complement of bitline  1 ). In practice, the number of wordlines and bitlines may be any number. 
   Data cell array  105  includes a first data cell  121  coupled between WL 1  and BT 0 , a second data cell  122  coupled between WL 0  and BC 0 , a third data cell  123  coupled between WL 1  and BT 1 , a fourth data cell  124  coupled between WL 0  and BC 1 , a fifth data cell  125  coupled between WL 3  and BT 0 , a sixth data cell  126  coupled between WL 2  and BC 0 , a seventh data cell  127  coupled between WL 3  and BT 1  and an eighth data cell  128  coupled between WL 2  and BC 1 . Each data cell  121  through  128  includes an NFET and a storage capacitor. The first source/drain of each NFET is coupled to that data cells respective bitline and the second source/drain of the NFET to a first plate of the capacitor. The second plate of each storage capacitor is coupled to ground and the gate of each NFET is coupled to that data cells respective wordline. 
   Reference cell array  110  includes a first reference cell  131 , a second reference cell  132 , a third reference cell  133  and a fourth reference cell  134 . The number of reference cells is equal to the number of bitlines in eDRAM array  100 . Reference cells  131  through  134  each include a first NFET, a second NFET and a storage capacitor. 
   For each reference cell  131  through  134 , the first source drain of each NFET and the first plate of the storage capacitor share a common node and the second plate of the storage capacitor is coupled to ground. 
   For second reference cell  132 , the second source/drain of the first NFET is coupled to BC 0  and the gate of the first NFET is coupled to a first reference wordline (RWL 0 ). The second source/drain of the second NFET is coupled to V REFX  and the gate of the second NFET is coupled to a first reference equalize signal line (REQ 0 ). 
   For first reference cell  131 , the second source/drain of the first NFET is coupled to BT 0  and the gate of the first NFET is coupled to a second reference wordline (RWL 1 ). The second source/drain of the second NFET is coupled to V REFX  and the gate of the second NFET is coupled to a second reference equalize signal line (REQ 1 ). 
   For fourth reference cell  134 , the second source/drain of the first NFET is coupled to BC 1  and the gate of the first NFET is coupled to RWL 0 . The second source/drain of the second NFET is coupled to V REFX  and the gate of the second NFET is coupled to REQ 0 . 
   For third reference cell  133 , the second source/drain of the first NFET is coupled to BT 1  and the gate of the first NFET is coupled to RWL 1 . The second source/drain of the second NFET is coupled to V REFX  and the gate of the second NFET is coupled to REQ 1 . 
   Reference cells transfer charge to the bitline of each bitline pair that is not being actively written to or read out. For example, if BT 0  is being read, then BC 0  is coupled to reference cell  132 . 
   The signal on REQ 0  is generated by inversion of a WL ODD signal by inverter  135 . The signal on RWL 0  is generated by inversion of the inverted WL ODD signal by inverter  136 . The signal on REQ 1  is generated by inversion of the a WL EVEN signal by inverter  137 . The RWL 1  signal is generated by inversion the inverted WL EVEN signal by inverter  138 . WL ODD is active when any odd numbered wordline is active, in this example WL 1  or WL 3 . WL EVEN is active when any even numbered wordline is active, in this example WL 0  or WL 2 . There are only two reference wordlines, RWL 1  representing odd numbered wordlines in data cell array  105  and RWL 0  representing even numbered wordlines in data cell array  105 . 
   Sense amplifier (SA) section  115  includes a first sense amplifier  141 , a second sense amplifier  142 , a first bitline restore circuit  151  and second bitline restore circuit  152 . The number of sense amplifiers and bitline restore circuits is equal to the number bitline pairs in eDRAM array  100 . 
   In the present example, sense amplifiers  141  and  142  are common cross-coupled amplifiers. Each sense amplifier  141  and  142  includes two PFET/NFET stacks, each PFET/NFET stack having a common node (where the drain of the NFET couples to the drain of the PFET). The gates of the NFET and the PFET of the each stack are coupled to each other and cross-coupled to the common node of the other stack. 
   For both first sense amplifier  141  and second sense amplifier  142 , the sources of the PFETS are coupled to a sense amplifier enable line (SETP) and the source of the NFETS are coupled to ground. For first sense amplifier  141 , the common node of the first NFET/PFET stack is coupled to BT 0  and the common node of the second NFET/PFET stack is coupled to BC 0 . For second sense amplifier  142 , the common node of the first NFET/PFET stack is coupled to BT 1  and the common node of the second NFET/PFET stack is coupled to BC 1 . 
   The signal on SETP is generated by transistors  160 A and  160 B in response to a sense amplifier enable signal (SETN) low which will bring SETP high or an equalize signal (EQ) high which will bring SETP low. (During equalization, the BT and BC lines of a bitline pair are shunted together.) With SETP low, there is no amplification of the signals on any of the bitlines. 
   First and second restore circuits  151  and  152  each include three NFETs in series, the gates of each NFET coupled to a signal line carrying EQ. For each restore circuit  151  and  152 , the source of the first NFET and source of the third NFET are coupled to ground. For first restore circuit  151 , a first source/drain of the second NFET is coupled to the drain of the first NFET and to BT 0 . A second source/drain of the second NFET is coupled to the drain of the third NFET and to BC 0 . For second restore circuit  152 , a first source/drain of the second NFET is coupled to the drain of the first NFET and to BT 1 . A second source/drain of the second NFET is coupled to the drain of the third NFET and to BC 1 . When EQ is high bitlines BT 0 , BC 0 , BT 1  and BC 1  are pulled low and the second NFET equalizes the true and complement of each bitline. 
   Column select circuits  120  are comprised of one NFET for each bitline, each NFET acting to gate the output of that bitline. In the present example, there are four NFETs  161 ,  162 ,  163  and  164 . A first source/drain of NFET  161  is coupled to BT 0  and a second source/drain of NFET  161  is coupled to an input/output node, data true (DT), of eDRAM array  100 . A first source/drain of NFET  162  is coupled to BC 0  and a second source/drain of NFET  162  is coupled to an input/output node, data compliment (DC), of eDRAM array  100 . The gates of NFETS  161  and  162  are coupled to a column select zero signal line (CS 0 ). A first source/drain of NFET  163  is coupled to BT 1  and a second source/drain of NFET  163  is coupled to DT. A first source/drain of NFET  164  is coupled to BC 1  and a second source/drain of NFET  164  is coupled to DC. The gates of NFETS  163  and  164  are coupled to a column select zero signal line (CS 1 ). There is one column select line for each bitline pair in eDRAM array  100 . Thus, NFETs  161 ,  162 ,  163  and  164  act as bit switches, gating input and output through nodes DT and DC. 
   An eDRAM is written or read by bringing a selected wordline high to transfer the charge stored in all cells coupled to that wordline to the bitlines. The sense amplifiers amplify this small amount of charge and the column select connect the amplified signals to read/write data path devices corresponding to the activated column select devices. The restore circuit precharges the bitlines after the sense amplifiers are turned off. 
     FIG. 2  is a timing diagram for the eDRAM of FIG.  1 . The timings sequence in eDRAM array  100  (see  FIG. 1 ) during the active portion of the cycle are (1) transfer the charge from the cell onto the bitline(s), (2) amplify the charge on the bitline(s) and (3) write back charge to the cell. The timings sequence in eDRAM array  100  (see  FIG. 1 ) during the restore (or precharge) portion of the cycle are (1) deselect wordlines; (2) turn off the sense amplifiers and (3) equalize the bitline(s) and precharge the bitlines(s) to GND before the start of the next read cycle. Since all timings are synchronized with the WL 0 / 2  or WL 1 / 3  signals, jitter (indicated the dashed lines) in WL 0 / 2  and WL 1 / 3  signals can shorten or lengthen the duration of the active period and the restore period. Jitter is the uncertainty in the timing (or clocking) of a signal edge. Referencing the CELL signal, if the active period is too short, then not enough charge will be written back to the cell and if the active period is too long, the EQ signal will not be long enough to bring both BC and BT to the same voltage value and a differential voltage will exist on the bitline pair that must be overcome by the sense amplifiers on the next cycle. 
     FIG. 3  is a block schematic diagram of a circuit for precise timing control of the start of the restore period of the eDRAM of  FIG. 1  according to a first embodiment of the present invention. In  FIG. 3 , a tunable timing circuit  200  includes a delayed lock loop (DLL) circuit  205  and a tunable delay circuit  210 . DLL circuit  205  includes a DLL controller  215 , a DLL filter  220 , a 360° delay element  225 , a clock-plus-one generator  230  and a comparator  235 . DLL controller  215 , delay element  225  and clock-plus-one generator  230  all receive a CLK signal  240  from an external source. DLL controller generates a control signal  245 , which is a word N-bits wide and is synchronous with CLK signal  240 . Control signal  245  is received by delay element  225  and LL filter  220 . DLL element  225  includes, in the present example, 18 delay stages (more or less stages may be used), each delay stage responsive to control signal  245 . Each stage is capable of adding to CLK signal  240  a 20° delay +/− an adjustable amount of delay responsive to control signal  245 . DLL element  225  is described in more detail infra with reference to FIG.  4 . The output of delay element  225  is coupled to a first input or comparator  235  and is a delayed CLK signal  250 , whose exact phase relative to the phase of CLK signal  240  is a function of control signal  245 . The output of clock-plus-one generator  230  is a shifted CLK signal  255 , which is shifted from CLK signal  240  by a whole phase or 360° and which is coupled to a second input of comparator  235 . Comparator  235  determines which of the phase of shifted CLK signal  255  or delayed CLK signal  250  is greater and generates a single-bit add/subtract delay signal  260 . Add/subtract delay signal  260  is coupled to DLL controller  215  and is used by the DLL controller to modify control signal  245  in order to add or remove adjustable delay from the delay stages of delay element  225 . DLL filter  220  filters control signal  245  to remove/reduce jitter in control signal  245  and generates a filtered control signal  265 . 
   Tunable delay circuit  210  includes, in the present example, a 320° delay element  270 A having 16 delay stages, a 280° delay element  270 B having 14 delay stages, a 240° delay element  270 C having 12 delay stages, a 200° delay element  270 D having 10 delay stages, a multiplexer  275  and a fine-tune delay circuit  280 . Except for the number of delay stages, delay elements  270 A to  270 D are similar to delay element  225 . The phase of a restore enable signal  285  generated by fine-tune delay circuit  280  and base delay on delay elements  270 A to  270 D. Delay elements  270 A to  270 D are coupled to CLK signal  240  and filtered control signal  265  and generate delayed CLK signals  290 A to  290 D, which are coupled to multiplexer  275 . In response to FUSE 1  control signals  295 A (which in the present example is a 4-bit word), multiplexer  275  selects one of delayed CLK signals  290 A to  290 D and couples the selected delayed CLK signal, designated course enable signal  290 S, to fine-tune delay circuit  280 . Fine-tune delay circuit  280 , in response to FUSE 2  control signals  295 B further tunes delayed CLK signal  290 S to generate restore enable signal  285 . 
   The following example illustrates the effect of tunable timing circuit  200 . Assume CLK signal  240  is a 100MHz (or 10 nano-seconds (ns)) signal. If delay signal  290 D is selected by multiplexer  275 , the phase course enable  290 S will be 200° out of phase from CLK signal  240  or lag behind by (200/360)×10=5.55 ns. If fine-tune delay circuit  280  adds a further 0.02 ns delay, then restore enable signal  285  will be delayed 5.57 ns relative to CLK signal  240 . 
   Returning to  FIG. 2 , restore enable signal  285  may be used to accurately and precisely time (gate) the turning on/off of wordline signals WL 0 / 2  and WL 1 / 3  of FIG.  2 . In other words, restore enable signal  285  can be used to very accurately time the start of the restore period of an eDRAM cycle since all signals, CELL, BT/BC EQ, RWL 0 , RWL 1 , WL 0 / 2  and WL 1 / 3  are synchronous and derived from the same CLK signal  240  (see FIG.  3 ). The present invention has the advantages that since DLL phase generators do not vary with process, voltage or temperature, thus changes in active/restore periods of the eDRAM cycle will also not vary with these parameters when driven by restore enable signal  285 . Control of active/restore cycles is not a function of CLK  240  duty cycle and further, jitter can be controlled to less than 50 pico-seconds (ps). 
     FIG. 4  is a schematic diagram of an exemplary delay element of the circuit of FIG. 3 . In  FIG. 4 , delay element  225  includes delay stages  300 A to  300 Q. In the present example, there are  18  delay stages. Each delay stage  300 A to  300 Q includes an inverter  305 , four capacitors  310 A to  310 D having respective capacitive values of C 1 , C 2 , C 3  and C 4  where C 1 &lt;C 2 &lt;C 3 &lt;C 4  and four corresponding NFETs  315 A to  315 D acting as pass gates. The drain of each NFET  315 A is coupled to a node  320  connecting the output of a previous inverter  305  to the input of a subsequent inverter. The source of NFET  315 A is coupled to GND through capacitor  310 A, the source of NFET  315 B is coupled to GND through capacitor  310 B, the source of NFET  315 C is coupled to GND through capacitor  310 C, The source of NFET  315 D is coupled to GND through capacitor  310 D. The gate of each NFET  315 A,  315 B,  315 C and  315 D is coupled to different respective bits N 0 , N 1 , N 2  and N 3  of control signal  245 . If a particular bit N 0  to N 3  is on (high), then the corresponding capacitor  310 A to  310 D is coupled to node  320 , increasing the delay through each stage  300 A to  300 Q. The primary delay through each stage  300 A to  300 Q is due to inverters  305 . In the present example, there are 2 4  possible delay settings possible. 
   If delay element  225  is nominally designed as a 360° phase delay element, then both positive and negative adjustment is possible by designing the nominal delay with one or more bits N 0  to N 4  expected to be on (high). Thus, some capacitive loading is included in the 20° phase shift of each stage  300 A to  300 Q and by turning off (bit low) particular bits, less than 20° phase delay will be realized through each delay stage (for the same clock cycle, voltage, process and temperature) and by turning on particular bits (bit high) more less than 20° phase delay will be realized through each delay stage. It should be pointed out 18 delay stages and four NFET/capacitor pairs per stage is exemplary and that the more delay stages and the more NFET/capacitor pairs (an bits in control signal  245 ) the more granularity of control will be realized. Delay elements  290 A to  290 D of  FIG. 3  are similar to delay element  225 , only the number of delay stages and the total delay through the respective delay elements being different. Because filtered control signal  265  (derived from control signal  245 , see  FIG. 3 ) is also four-bits, then there are four NFET/capacitor pairs in each stage of each delay element  290 A to  290 D. 
     FIG. 5  is a schematic diagram of an exemplary fine-tuning circuit of the circuit of FIG.  3 . In  FIG. 5 , fine-tune delay circuit  280  includes four inverter banks  325 A,  325 B,  325 C and  325 D, each inverter bank coupled between course enable signal  290 S and a de-multiplexer  330 . The output of multiplexer  330  is restore enable signal  285 . Multiplexer  330  is responsive to FUSE 2  control signals  295 B. Inverter bank  325 A comprises two inverters | 1  and | 2  connected in series, inverter bank  325 B comprises four inverters | 3 , | 4 , | 5  and | 6  connected in series, inverter bank  325 C comprises six inverters | 7 , | 8 , | 9 , | 10 , | 11  and | 12  connected in series and inverter bank  325 D comprises eight inverters | 13 , | 14 , | 15 , | 16 , | 17 , | 18 , | 19  and | 20  connected in series. By selection of various combinations of inverter banks  325 A to  325 D, small increases or decreases to the phase of course enable signal  290 S are realized in restore enable signal  285 . There are four inverter banks  325 A,  325 B,  325 C and  325 D because FUSE 2  control signal  295 B is a four-bit control word. More or less bits and a corresponding number of inverter banks may be used. 
     FIG. 6  is a schematic diagram of an exemplary fuse bank circuit  335  for generating control signals for the circuit of FIG.  3 . In  FIG. 6 , fuse bank circuit  335  includes a four-bit shift register  340  having a first stage  345 A, a second stage  345 B, a third stage  345 C and a fourth stage  345 D, a first fuses  350 A, a second fuse  350 B, a third fuse  350 C and a fourth fuse  350 D and a first multiplexer  355 A, a second multiplexer  355 B, a third multiplexer  355 C and a fourth multiplexer  355 D. First stage  345 A of shift register  340  and first fuse  350 A are coupled to inputs of first multiplexer  355 A. The output of first multiplexer  355 A is an F 0  bit of FUSE 1  (or FUSE 2 ) control signal  295 A (or  295 B). Second stage  345 B of shift register  340  and second fuse  350 B are coupled to inputs of second multiplexer  355 B. The output of the second multiplexer  355 B is an F 1  bit of FUSE 1  (or FUSE 2 ) control signal  295 A (or  295 B). Third stage  345 C of shift register  340  and third fuse  350 C are coupled to inputs of third multiplexer  355 C. The output of third multiplexer  345 C is an F 2  bit of FUSE 1  (or FUSE 2 ) control signal  295 A (or  295 B). Fourth stage  345 D of shift register  340  and fourth fuse  350 D are coupled to inputs of fourth multiplexer  355 D. The output of fourth multiplexer  355 D is an F 3  bit of FUSE 1  (or FUSE 2 ) control signal  295 A (or  295 B). Shift register  340  is a typical level sensitive scan design (LSSD) register. 
   The pattern of fuses  350 A,  350 B,  350 C and  350 D to blow is determined during test of the eDRAM  100  (see FIG.  1 ). A test pattern is scanned into shift register  340  to simulate a fuse blow pattern and the performance or other parameters of the eDRAM are measured. For example, the tester could determine which pattern gives the highest yield or which pattern gives the fastest eDRAM. The fuses are then blown to this pattern. Each eDRAM on a single chip on a wafer may be individually tested and appropriate fuses blown. Alternatively, a representative number of eDRAMs on several chips on a wafer may be tested, and an “average” pattern selected for blowing the same fuses on all eDRAMs on all chips. Or a representative number of eDRAMs on several chips on several wafers may be tested, and an “average” pattern selected for blowing the same fuses on all eDRAMs on all chips on all wafers in a lot (group of wafers processed together). 
   Fuses  350 A,  350 B,  350 C and  350 D may be laser blow fuses, electrical blow fuses or electrical blow antifuses. More or less than four fuses may be used depending upon the number of delay elements in tunable delay circuit  210  (see  FIG. 3 ) and inverter banks in fine-tune delay circuit  280  (see FIG.  5 ). 
   Optionally, two separate fuse bank circuits  335  may be replaced with a single fuse bank circuit having an eight-bit shift register with two banks of four fuses each in order to share a scan chain. 
   The present invention is extendable beyond control of just the restore (precharge) period of an eDRAM to any or all of the synchronous control signals or functions of an eDRAM describes supra. 
     FIG. 7  is a block schematic diagram of an eDRAM  400 A according to a second embodiment of the present invention. In  FIG. 7 , eDRAM  400 A includes a latch receiving an address signal  405  and CLK signal  240 . Address signal  405  is received by a latch  410  and decoded by a decoder  415  which generates a wordline select signal  420  received by a wordline driver  425  which generates a wordline signal  430  received by a memory array  440 . CLK signal  240  is also received by DLL circuit  205 , which generates a filtered control signal  265 A. Filtered control signal  265 A is similar to filtered control signal  265  (see  FIG. 3 ) except that filtered control signal  265 A is an n-bit word, where n is any positive whole number. Filtered control signal  265 A is received by a first, a second, a third and a fourth delay elements  445 ,  450 ,  455  and  460  which generate a SA set enable signal  465 , a CS enable signal  470 , a BL restore enable signal  475  and an output enable signal  480  respectively. First, second, third and fourth delay elements  445 ,  450 ,  455  and  460  are similar to delay elements  290 A to  290 D illustrated in FIG.  3  and described supra, except the phase shifts are different. 
   In the present example, first delay element  445  has a 180° delay, second delay element  450  has a 190° delay, third delay element  455  has a 280° delay and fourth delay element  460  has a 340° delay. First delay element  445  has a lesser delay than second delay element  450 , which has a lesser delay than third delay element  455 , which has a lesser delay than fourth delay element  460 , otherwise eDRAM  400  will not function properly. In the second embodiment of the present invention the delays of delay elements  445 ,  450 ,  455  and  460  are fixed and “designed in”. SA set enable signal  465  is received by sense amplifier control circuit  485 . CS enable signal  470  is received by CS control circuit  490 . BL restore enable signal  475  is received by WL driver circuit  425 , SA control circuit  485  and BL restore control circuit  495 . Output enable signal  480  is received by and output latch  500 . SA control circuit  485  generates a SA set signal  505  received by memory array  440 . BL restore control circuit  495  generates a BL restore signal (EQ signal)  510  received by memory array  440 . CS control circuit  490  generates a CS signal  515  received by memory array  440 . Output latch  500  receives data  520  from memory array  440  and sends the data to OCD  525 . 
     FIG. 8  is a block schematic diagram of an eDRAM  400 B according to a third embodiment of the present invention. In  FIG. 8 , eDRAM  400 B is similar to eDRAM  400 A of  FIG. 7 , except that first, second, third and fourth fixed delay elements  445 ,  450 ,  455  and  460  of eDRAM  400 A are replaced respectively by first, second, third and fourth tunable delay circuits  530 ,  535 ,  540  and  545  respectively in eDRAM  400 B. First, second, third and fourth tunable delay circuits  530 ,  535 ,  540  and  545  are similar to tunable delay circuit  210  illustrated in FIG.  3  and described supra. The delay of first tunable delay circuit  530  is “programmed” via signals FUSEA and FUSEB. The delay of second tunable delay circuit  535  is “programmed” via signals FUSEC and FUSED. The delay of third tunable delay circuit  540  is “programmed” via signals FUSEE and FUSEF. The delay of fourth tunable delay circuit  545  is “programmed” via signals FUSEG and FUSEH. 
   The description of the embodiments of the present invention is given above for the understanding of the present invention. It will be understood that the invention is not limited to the particular embodiments described herein, but is capable of various modifications, rearrangements and substitutions as will now become apparent to those skilled in the art without departing from the scope of the invention. Therefore, it is intended that the following claims cover all such modifications and changes as fall within the true spirit and scope of the invention.