Abstract:
Methods and apparatus for determining a dominant frequency contained in a multi-frequency analog signal. An apparatus includes a first matched filter pair and a second a second matched filter pair. Both the first matched filter pair and the second matched filter pair are configured to receive in-phase (I-phase) and quadrature-phase (Q-phase) digital samples of a digitized multi-frequency analog signal. The summed magnitudes of outputs of the first and second matched filter pairs provide an indication of a dominant frequency represented in the digital samples. The first matched filter pair may include first and second matched filters, where the first matched filter includes a multiplier operable to multiply, on a sample-by-sample basis, a sample by a coefficient. A sample comprises an n-bit digital representation of either the I-phase amplitude or the Q-phase amplitude at a given instant. The coefficients are the terms required to complete the matching functions (e.g. 1-bit Walsh coefficients).

Description:
FIELD OF THE INVENTION 
     The present invention relates to electronic systems and methods. More particularly, the present invention relates to systems and methods for determining a dominant frequency contained in analog signals having multiple frequencies. 
     BACKGROUND OF THE INVENTION 
     A transmitter in a radio frequency (RF) communications system modulates an electromagnetic wave carrier signal by impressing information (e.g. voice, image, data, etc.) on a carrier wave having a frequency that can be propagated over the airwaves. In the case of a digital communications system, the information is in the form of a stream of data bits, where each data bit has either a value of “0” or a value of “1”. One commonly used modulation technique is frequency shift keying (FSK). The FSK technique operates by shifting a continuous carrier frequency in a binary manner to either one or the other of two discrete frequencies. One frequency is designated as the “mark” frequency and the other as the “space” frequency. The mark and space frequencies correspond to a binary “1” and a binary “0”, respectively. This FSK modulation scheme (also known as “binary FSK”) is shown in  FIG. 1 , where the space frequency is indicated by a first frequency shifted carrier of frequency f 1 , and the mark frequency is indicated by a second frequency shifted carrier frequency of frequency f 2 . 
     One particular and popular use of FSK is in radio frequency identification (RFID) systems. Among other applications, RFID systems are used for inventory control, supply chain management, and anti-theft of merchandise in stores. A typical RFID system  20  is shown in  FIG. 2 . RFID system  20  comprises a plurality of transponders (referred to in the art as “tags”)  200  and one or more transceivers (referred to in the art as “interrogators” or “readers”)  202 . A reader  202  includes an antenna  204 , which allows it to interrogate one or more of the tags  200  over a forward wireless link. The tags  200  also have their own respective antennas  208 , which allow them to transmit tag information back to the reader  202  over reverse wireless link. The reader  202  and other readers (not shown in  FIG. 2 ) communicate with a host computer  210 . Data collected from the tags  200  can either be sent directly to the host computer  210  through standard interfaces, or it can be stored in the reader  202  and later uploaded to the host computer  210 , either directly or by a wireless link, for data processing. 
     Tags are typically embodied as semiconductor microchips having small amounts of memory for storing the tag&#39;s ID number and, in some applications, information concerning the item to which the tag is associated. Further, tags are either “passive” or “active”, depending on how they are powered. Active tags contain their own on-board power source, i.e. a battery, which the active tag uses to process received signals and to transmit tag information back to a reader. Passive tags do not have batteries. They derive their energy from RF signals broadcast by the reader and electromagnetically coupled to the tag antennae. Part of the coupled electromagnetic energy is rectified and stored in each tag. Passive tags use this stored energy as a power source to operate the logic and the RF modulator so as to send data back to the reader by a technique known as backscatter modulation. 
     In order for the reader  202  to address any particular tag (i.e. Tag A, B, C, D or E) from the population of tags, a process known as “singulation” is typically used. To singulate a tag from the population of tags, the reader  202  polls the tags  200  for their ID numbers (or derivative thereof), typically on a bit-by-bit basis. Because multiple tag responses may interfere with one another, anti-collision algorithms are typically employed in the singulation process. Anti-collision algorithms are either probabilistic or deterministic. One well-known probabilistic anti-collision algorithm is the Aloha technique, whereby tags respond to a polling signal from the reader  202  at random intervals. If a collision occurs, the tags responsible for the collision wait for another, usually longer, time interval before responding again. A known deterministic anti-collision algorithm is the so-called “binary tree-walking” algorithm. According to this approach, the reader  202  initially polls the tags  200  for the first bit of the tags&#39; respective ID numbers. Based on the bit values received, the reader  202  then limits the number of tags which are to send subsequent bits of their ID numbers. This process is repeated until the ID of a single tag has been singulated. 
     In an FSK RFID system success of the anti-collision algorithm is conditioned upon the reader being capable of discriminating between the two FSK frequencies employed to represent binary “0&#39;s” and binary “1&#39;s, both of which may be received at the same time. It would be desirable, therefore, to have a frequency determining apparatus and method capable of determining a dominant frequency contained in simultaneously received signals having multiple frequencies. 
     SUMMARY OF THE INVENTION 
     Methods and apparatus for determining a dominant frequency contained in analog signals having multiple frequencies are disclosed. According to an exemplary aspect of the invention, an apparatus for determining a dominant frequency in digital samples of a multi-frequency analog signal includes a first matched filter pair and a second a second matched filter pair. Both the first matched filter pair and the second matched filter pair are configured to receive in-phase (I-phase) and quadrature-phase (Q-phase) digital samples of a digitized multi-frequency analog signal, where the I and Q are referred to as being in quadrature, or otherwise known as orthogonal or independent. The summed magnitudes of outputs of the first and second matched filter pairs provide an indication of a dominant frequency represented in the digital samples. The first matched filter pair may include first and second matched filters, where the first matched filter includes a multiplier operable to multiply, on a sample-by-sample basis, a sample by a coefficient. A sample comprises an n-bit digital representation of either the I-phase amplitude or the Q-phase amplitude at a given instant. The coefficients are the terms required to complete the matching functions (e.g. 1-bit Walsh coefficients). 
     According to another aspect of the invention, a method of determining an amplitude of a dominant frequency represented in digital samples of a multi-frequency analog signal includes sampling a multi-frequency analog signal at a sampling rate to generate digital samples; multiplying, on a sample-by-sample basis, samples by a first set of coefficients associated with a first frequency possibly corresponding to the dominant frequency; multiplying, on a sample-by-sample basis, samples by a second set of coefficients associated with a second frequency possibly corresponding to the dominant frequency; and using the results of multiplying to determine the dominant frequency represented in the digital samples. 
     According to another aspect of the invention, a digital frequency determining apparatus comprises a radio frequency (RF) receiver configured to receive a multi-frequency RF signal and convert it to a baseband signal having an in-phase (I-phase) component and a quadrature-phase (Q-phase) component; an analog-to-digital converter (ADC) operable to sample the I-phase and Q-phase baseband signal components to produce I-phase and Q-phase digital samples; a first matched filter pair configured to receive said I-phase and Q-phase digital samples; and a second matched filter pair configured to receive said I-phase and Q-phase digital samples. The summed magnitudes of outputs of said first and second matched filter pairs provide an indication of a dominant frequency present in the multi-frequency RF signal. 
     The summary of the aspects of the invention described above are meant only to provide a few aspects of the invention. Other aspects of the invention are described in the detailed description of the invention below and the claims set forth at the end of this disclosure. A further understanding of the scope, nature and advantages of the invention may be realized by reference to the remaining portions of the specification and the attached drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  illustrates the frequency shift keying (FSK) modulation scheme known in the prior art; 
         FIG. 2  shows a typical RFID system known in the prior art; 
         FIG. 3  shows a digital frequency determining apparatus, according to an embodiment of the present invention; 
         FIG. 4  shows an orthogonal set of Walsh functions having 1-bit coefficients sampled in time, which may be used in the various matched filters of the digital frequency determining apparatus in  FIG. 3 , according to an embodiment of the present invention; 
         FIG. 5  shows an exemplary FSK receiver employing the digital frequency determining apparatus in  FIG. 3 , according to an embodiment of the present invention; 
         FIG. 6  shows an exemplary FSK receiver employing the digital frequency determining apparatus in  FIG. 3 , wherein the digital frequency determining apparatus is implemented in an FPGA, according to an embodiment of the present invention; and 
         FIG. 7  shows an exemplary FSK RFID system having a reader employing the digital frequency determining apparatus in  FIG. 3 , according to an embodiment of the present invention. 
     
    
    
     DETAILED DESCRIPTION 
     Embodiments of the present invention include frequency determining apparatus and methods capable of determining a dominant frequency contained in simultaneously received signals having multiple frequencies. Those of ordinary skill in the art will realize that the following detailed description of the present invention is illustrative only and is not intended to be in any way limited to a particular exemplary embodiment. Other embodiments of the invention will readily suggest themselves to such skilled persons having the benefit of this disclosure. Reference will now be made in detail to implementations of the present invention as illustrated in the accompanying drawings. The same reference indicators will be used throughout the drawings and the following detailed description to refer to the same or similar parts. 
     Referring to  FIG. 3 , there is shown a digital frequency determining apparatus  30 , according to an embodiment of the present invention. As described in more detail below, the simplicity of the design and the type of components employed render the apparatus particularly suitable for implementation in a programmable device such as a field programmable gate array (FPGA). Apparatus  30  is coupled to a first digital auto-zero high-pass filter  300 , which is configured to receive an in-phase (I-phase) digital data stream (i.e. sequence of samples) from an I-channel input, and to a second digital auto-zero high-pass filter  302 , which is configured to receive a quadrature-phase (Q-phase) digital data stream from a Q-channel input. The digital auto-zero high-pass filters  300  and  302  operate to remove any DC offsets present in the I-phase and Q-phase digital data streams. 
     According to an aspect of the invention, the I and Q-phase digital data streams may be comprised of a predetermined number of digital data samples representing information received from multiple sources having multiple carrier frequencies (e.g. two subcarrier frequencies f 1  and f 2 ). The digitized data samples may represent, for example, data bits received from a plurality of tags of an FSK RFID system. Subcarrier frequencies f 1  and f 2  may be, for example, backscatter modulated subcarrier frequencies (e.g. mark and space frequencies) associated with an FSK RFID system. 
     The outputs of the digital auto-zero high-pass filters  300  and  302  are coupled to inputs of four matched filters  304 - 1 ,  304 - 2 ,  304 - 3  and  304 - 4 . In particular, the output of digital auto-zero high-pass filter  300  is coupled to the inputs of matched filters  304 - 1  and  304 - 3 , and the output of digital auto-zero high-pass filter  302  is coupled to the inputs of matched filters  304 - 2  and  304 - 4 . Each of the matched filters  304 - 1 ,  304 - 2 ,  304 - 3  and  304 - 4  is comprised of an I-channel and Q-channel that operate on the subcarriers of an FSK RFID system, for example. Each of the I-channels of the matched filters  304 - 1 ,  304 - 2 ,  304 - 3  and  304 - 4  includes an integrator (i.e. an “accumulator”)  306 , and an absolute value operator  308 . Similarly, each of the Q-channels of the matched filters  304 - 1 ,  304 - 2 ,  304 - 3  and  304 - 4  includes an integrator  310  and an absolute value operator  312  (or square law detector or equivalent). The I-channels of matched filters  304 - 1  and  304 - 2  also include 1-bit (e.g. 2&#39;s complement) multipliers  314  and  316 , which have first inputs configured to receive the filtered I-phase and Q-phase digital data streams from auto-zero high-pass filters  300  and  302 , respectively, and second inputs configured to receive a stream of 1-bit coefficients W 1 . The term 1-bit multiplier refers to a multiplier that multiplies a 1-bit coefficient by an n-bit sample. The W 1  coefficients may be generated, for example, by sampling a periodic signal (e.g. a sine wave) of a first frequency (e.g. a first subcarrier frequency f, associated with an RFID system) at a rate substantially equal to a sampling rate of an analog-to-digital (ADC) converter used to digitize the digital data streams. Similarly, the I-channels of correlators  304 - 3  and  304 - 4  also include 1-bit multipliers  318  and  320 , which have first inputs configured to receive the filtered in-phase and quadrature-phase digital data streams from auto-zero high-pass filters  300  and  302 , respectively, and second inputs configured to receive a stream of 1-bit coefficients W 2 . The W 2  coefficients may be generated, for example, by sampling a periodic signal (e.g. a sine wave) of a second frequency (e.g. a second subcarrier frequency f 2  associated with an RFID system) at a rate substantially equal to the sampling rate of an analog-to-digital (ADC) converter used to digitize the digital data streams. The 1-bit multipliers operate to multiply the value of each sample of the data streams by +1 or −1 depending on the condition of the particular coefficient (i.e. 1=+1 and 0=−1). 
     Similar to the I-channels described above, the Q-channels of matched filters  304 - 1  and  304 - 2  include 1-bit multipliers  322  and  324 , which have first inputs configured to receive the filtered I-phase and Q-phase digital data streams from auto-zero high-pass filters  300  and  302 , respectively, and second inputs configured to receive a stream of 1-bit coefficients W 1  but ninety-degrees out of phase to (i.e. in quadrature to) those received at the first inputs of multipliers  314  and  316  (indicated as W 1 +90° in  FIG. 3 ). Similarly, the Q-channels of matched filters  304 - 3  and  304 - 4  include 1-bit multipliers  326  and  328 , which have first inputs configured to receive the filtered in-phase and quadrature-phase digital data streams from auto-zero high-pass filters  300  and  302 , respectively, and second inputs configured to receive a stream of 1-bit coefficients W 2  but ninety-degrees out of phase to those received at the first inputs of multipliers  318  and  320  (indicated as W 2 +90° in  FIG. 3 ). The ninety-degree phase shift ensures that Wi and Wi+90° are orthogonal and therefore acquires all components of the incident signal for Wi. 
     As shown in  FIG. 3 , outputs of the absolute value operators  308  and  312  in each matched filter  304 - 1 ,  304 - 2 ,  304 - 3  and  304 - 4  are coupled to first and second inputs of intermediate summers  330 ,  332 ,  334  and  336 . Outputs of intermediate summers  330  and  332  are in turn coupled to inputs of output summers  338  and  340 , which provide output signals M(f 1 ) and M(f 2 ), respectively. Alternatively, the outputs of the absolute value operators  308  and  312  of matched filters  304 - 1  and  304 - 2  may be coupled to first and second multiple-input summers, to provide the output signals M(f 1 ) and M(f 2 ). 
     Each of the streams of 1-bit coefficients W 1  and W 2  are comprised of coefficients of value −1 or +1 (or negative and positive values of some sort). Preferably, both W 1  and W 2  are zero-mean functions, meaning that each has an equal number of coefficients of values −1 and +1. An equal number of −1 and +1 coefficients ensures that DC biases are not introduced in the matched filter operations. W 1  and W 2  may comprise Walsh functions, such as shown in  FIG. 4 , or may comprise other orthogonal functions having similar characteristics. 
     During operation, the digital frequency determining apparatus  30  shown in  FIG. 3  provides output signals M(f 1 ) and M(f 2 ). The larger of M(f 1 ) and M(f 2 ) over a given predetermined time period, signifies which of the carrier (e.g. subcarrier) frequencies f 1  and f 2  was most strongly represented in the stream of I-channel and Q-channel digital data streams during the given time period. In particular, each of the matched filters  304 - 1 ,  304 - 2 ,  304 - 3  and  304 - 4  receives the I-phase and Q-phase digital data streams. Then, for each bit received, the 1-bit multipliers  314 - 328  multiply, on a sample-by-sample basis, samples by 1-bit coefficients provided in functions W 1  and W 2 . For each multiplication, each multiplier  314 - 328  provides an output state having an absolute value greater than zero if the input signal bit and 1-bit coefficient have the same value (i.e. if they “match” or, as some skilled in the art refer to as “correlate”), and provides an output state having a zero value if the bits do not match (a signal in quadrature will not match or correlate). The integrators  306  and  310  in each of the I and Q paths of the matched filters  304 - 1 ,  304 - 2 ,  304 - 3  and  304 - 4  accumulate the results of the n-bit sample by 1-bit coefficient multiplications in their respective paths over the predetermined time period and provide accumulated results to the absolute value operators  308  and  312 , which convert any negative accumulated result to a positive value. The outputs of the absolute value operators  308  and  312  of matched filters  304 - 1  and  304 - 2  are summed by intermediate summers  330  and  332 , the results of which are summed by the output summer  338 . Similarly, the outputs of the absolute value operators  308  and  312  of correlators  304 - 3  and  304 - 4  are summed by intermediate summers  334  and  336 , the results of which are summed by the output summer  340 . The dominant frequency may then be determined by direct comparison of the magnitudes of output signals M(f 1 ) and M(f 2 ). The larger magnitude of M(f 1 ) and M(f 2 ) signifies the dominant frequency. 
     In addition to the foregoing, a minimum threshold can be set to ensure that the response of the digital frequency determining apparatus  30  is distinguishable from noise. Further, the magnitudes of M(f 1 ) and M(f 2 ), which provide an indication of signal strength, may be used to determine range or tag performance, for example, when the digital frequency determining apparatus  30  is used in an RFID application. 
     Those of ordinary skill in the art will readily understand and appreciate that the inventions described herein are not limited to using Walsh functions, and that other orthogonal functions may be used. Further, whereas the frequency determining apparatus  30  in  FIG. 3  is shown as determining the dominant one of two frequencies f 1  and f 2  (e.g. FSK RFID subcarrier frequencies), those of ordinary skill in the art will readily appreciate and understand that the digital frequency determining apparatus  30  may be generalized (i.e. extended) so that it can determine from among n frequencies (i.e. f 1 , f 2 , . . . , f n ), where n is an integer greater than or equal to two. 
     According to an embodiment of the present invention, the digital frequency determining apparatus  30  may comprise part of an FSK receiver  50 , as shown in  FIG. 5 . An antenna  500  of FSK receiver  50  is configured to simultaneously receive a signal having two frequency shifted subcarrier frequencies (i.e. FSK mark and space frequencies), one having a first frequency shift of f 1  (e.g. 2.2 MHz) and the other having a second frequency shift of f 2  (e.g. 3.3 MHz). A low-noise amplifier  502  amplifies the received signal and directs it to both an I-phase mixer  504  and a Q-phase mixer  506 . A local oscillator (LO) is coupled to an LO input of I-phase mixer  504 , and a ninety-degree phase-shifted version of LO is coupled to an LO input of Q-phase mixer  506 . The I-phase mixer  504  and Q-phase mixer  506  operate to down-convert the received signal to an intermediate frequency (IF). Outputs of the mixers  504  and  506  are coupled to inputs of switched capacitor auto-zeros  508  and  510 , respectively. Until the receiver  50  is ready to receive data, the switches of the switched capacitor auto-zeros  508  and  510  remain closed. Accordingly, during this time, whatever the output voltage the mixers  504  and  506  have is stored on the respective capacitors of the switched capacitor auto-zeros  508  and  510 . The switches of the switched capacitor auto-zeros  508  and  510  are maintained in their closed positions (i.e. remain shorted to ground) for a short time prior to edge transitions of the received signal, after which the switches are opened. In other words, the opening and closing of the switches of the switched capacitor auto-zeros  508  and  510  are timed so that they nearly follow the switching frequency of the transmitter, but are offset to a small degree so that any spikes can be effectively removed prior to edge transitions. This ensures that large voltage spikes on the received signal edges are not transmitted through the remainder of the receiver  50 . Outputs of the switched-capacitor auto-zeros  508  and  510  are coupled to inputs of low-pass filters (LPFs)  512  and  514 . The LPFs  512  and  514  have predetermined cutoff frequencies (e.g. 3.5 MHz), and are used to filter out noise and unwanted frequency byproducts generated by mixers  504  and  506 . Outputs of LPFs  512  and  514  are coupled to inputs of amplifiers  516  and  518 , respectively. The amplifiers  516  and  518  condition and amplify the filtered signals and couple them to high-pass filters (HPFs)  520  and  522 . The HPFs  520  and  522  serve to remove any DC offsets present in the signals received from the amplifiers  516  and  518 . Outputs of the HPFs  520  and  522  are coupled to a dual-channel analog-to-digital (ADC) converter  524 , which operates to quantize the incoming signals at a sampling rate and provide digital data samples. The dual-channel digital data samples are coupled to the inputs of the digital frequency determining apparatus  30  (labeled “DFDA” in the drawing), and, optionally, first through digital auto-zero high-pass filters (e.g. digital auto-zero high-pass filters  300  and  302 , as shown in  FIG. 3 ). 
     According to an embodiment of the present invention, the digital frequency determining apparatus  30  (with or without the auto-zero filters) may be implemented in a programmable device such as field-programmable gate array (FPGA). An FPGA implementation  60  is shown in  FIG. 6 . Although a microprocessor may be used to perform each of the various operations (e.g. multiply, integrate, absolute value, etc.) of the matched filters  304 - 1 ,  304 - 2 ,  304 - 3  and  304 - 4  sequentially, an FPGA can be advantageously configured so that it can perform many or all of the same operations (e.g. all of the multiplications, all of the integrations, etc.) simultaneously (i.e. in parallel). 
     The ability of the frequency determining apparatus  30  to determine the most dominant of two frequencies contained in multiple-frequency signals at the same time, makes it well suited for use in an FSK RFID system, where a plurality of tags may be transmitting “1&#39;s” represented by a first frequency shift f 1  and “0&#39;s” represented by a second frequency shift f 2  at the same time. For example, in an FSK RFID system, “0&#39;s” and “1&#39;s” are signaled by first and second tones (i.e. mark and space) for a plurality (e.g.  20 ) of system clock cycles. The frequency determining apparatus  30  can be employed to determine which of the tones has the largest magnitude, and, therefore, whether most of the tags transmitted a “0” or transmitted a “1”. The RFID system can then use this magnitude information to instruct each of the plurality of tags to either enter a suspended state (i.e. a state in which they will not respond to a next command by the reader) or to remain in a ready state in which their next bit is transmitted in response to the next command of the reader. This process can be repeated until a single tag has been singulated from the plurality of tags. 
       FIG. 7  shows an exemplary embodiment of an FSK RFID system  70 , which can perform the operations described above, and which utilizes the digital frequency determining apparatus  30 , also described above. The FSK RFID system  70  includes a reader  700  and a plurality of tags  702 - 1 ,  702 - 2 , . . . ,  702 -N. The reader  700  includes an antenna  704 ; a system clock  706  for controlling a receiver (RX)  708  and a transmitter (TX)  710 ; an ADC  712 ; the digital frequency determining apparatus  30 ; a coefficient memory or generator  713 ; a baseband processor  714 ; a memory  716 ; a DAC  718 ; a power amplifier (PA)  720 ; and an TX/RX switch  722  for coupling the antenna  704  to either the front-end of the receiver  708 , during times the reader  700  is receiving data, or, alternatively, to the output of the PA  720 , during times when the reader  700  is transmitting data. The Walsh function (or other zero-mean function) coefficients can be generated, for example, by sampling one or more sinusoidal waves at the desired coefficient stream frequencies; by utilizing one or more frequency synthesizers; by employing a rate generator, by deriving the coefficients from one or more state machines; by retrieving the coefficients from a look up table (LUT), etc. Using one or more state machines is beneficial since the state machines and some, or the remainder of, the digital frequency determining apparatus  30  elements can be implemented in an FPGA or other programmable device, as described above. 
     While particular embodiments of the present invention have been shown and described, it will be obvious to those skilled in the art that, based upon the teachings herein, changes and modifications may be made without departing from this invention and its broader aspects. For example, whereas the digital frequency determining apparatus  30  is shown and described in the context of binary FSK, those of ordinary skill in the art will readily understand and appreciate that the principles of the present invention may be extended to M-ary FSK implementations by simply adding additional matched filters and providing additional coefficient generators. Further, those of ordinary skill in the art can readily appreciate and understand that the digital frequency determining apparatus  30  may be modified so that it can operate using QAM (quadrature amplitude modulation), by replacing the 1-bit multipliers with appropriate higher order multipliers. Still further, whereas a specific exemplary embodiment of the invention is described in the context of an exemplary RFID system  70 , those of ordinary skill in the art will readily appreciate and understand that the concepts underlying the present invention may be applied to other frequency determining contexts requiring the determination of a frequency contained in simultaneously received signals having multiple frequencies. Finally, whereas the exemplary embodiments have been described as using traditional tonal components, quadrature non-tonal signals can also be used. Therefore, the appended claims are intended to encompass within their scope all such changes and modifications as are within the true spirit and scope of this invention.