Abstract:
A direct conversion receiver (DCR) calibrated for phase and gain mismatch is provided. The DCR comprises a poly-phase filter that generates mismatched in-phase and quadrature-phase differential signals; the in-phase differential signal is mixed with first and second local oscillation signals; the quadrature-phase differential signal is mixed with third and fourth local oscillation signals; the first and second local oscillation signals have a first adjustable phase mismatch, and the third and fourth oscillation signals have a second adjustable phase mismatch; A mismatch estimation unit (MEU) estimates the entire phase/gain mismatch (signal distortion) of the DCR, The first and second adjustable phase mismatches are adjusted so that the signal distortion of the DCR, as estimated by the MEU is minimized. Thereafter, reduced-component DCRs (without an MEU), calibrated for the mismatches of a poly-phase filter, may be mass produced.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    The present invention relates to a radio frequency (RF) signal receiver, and more particularly, to a direct conversion RF signal receiver.  
           [0003]    2. Description of the Related Art  
           [0004]    A direct conversion method is a method by which a signal in an RF band is down-converted into a baseband signal by mixing once the RF signal with a local oscillation signal. That is, the RF band signal is directly converted into a baseband signal, not converted into an intermediate frequency (IF) band signal and then into a baseband signal. In general, the former is referred to as a heterodyne method, while the latter is referred to as a direct conversion method.  
           [0005]    [0005]FIG. 7 is a diagram of a conventional direct conversion receiver (hereinafter referred to as a DCR). When the conventional DCR receives an RF signal, a low noise amplifier  110  amplifies the RF signal and a transformer  120  converts the amplified RF signal into a differential signal. The differential signal is converted into an in-phase differential signal (I_W RF ) and a quadrature-phase differential signal (Q_W RF ) through a poly-phase filter  130  and input to a down-conversion mixer unit  710 . The mixer unit  710  comprises a first mixer  711  and a second mixer  712  that mix the in-phase differential signal (I_W RF ) with a first local oscillation signal (OS 1 ) and a second local oscillation signal (OS 2 ), respectively; a third mixer  713  and a fourth mixer  714  that mix the quadrature-phase differential signal (Q_W RF ) with the first and second oscillation signals (OS 1 , OS 2 ), respectively; and filters  721  through  724  that low pass filter the outputs of the respective mixers  711  through  714 . Here, the second local oscillation signal (OS 2 ) has the same oscillation frequency as that of the first local oscillation signal (OS 1 ), but a 90° phase difference from the first local oscillation signal (OS 1 ).  
           [0006]    A subtracter subtracts the output signal (QQ) of the fourth mixer  714  from the output signal (II) of the first mixer  711  to output an I-path signal (I_PATH). An adder adds the output signal (IQ) of the second mixer  712  and the output signal (QI) of the third mixer  713  to output a Q-path signal (Q_PATH). The I-path signal (I_PATH) and Q-path signal (Q_PATH) are baseband signals converted from an RF signal.  
           [0007]    The DCR having the structure shown in FIG. 7 usually has a phase and gain mismatch. The phase and gain mismatch in the DCR occurs in the poly-phase filter  130  and the mixer unit  710 . Ideally the phase difference of the in-phase differential signal (I_W RF ) and the quadrature-phase differential signal (Q_W RF ) output from the poly-phase filter  130  is 90°, but the actual phase difference is not. Also, ideally, the phase difference of the first and second local oscillation signals (OS 1 , OS 2 ) provided to the mixers  711  through  714  is 90° and the gains of the output signals of the mixers  711  through  714  are the same. However, the phase difference of the first and second location oscillation signals (OS 1 , OS 2 ) is actually  90 ±φ (causing a phase mismatch. Also, a gain mismatch occurs in the output signals of the mixers.  
           [0008]    If the phase and gain mismatch occurs in the DCR, as described above, the error rate of the received signal increases due to the mismatch, or the signal fidelity is degraded. Accordingly, to prevent distortion of a signal and to obtain a desired signal, it is important to identify the degree of phase and gain mismatch degree in the DCR and to calibrate for the mismatch.  
           [0009]    However, to solve the phase mismatch the conventional methods have focused on a local oscillator, which generates a local oscillation signal. That is, most efforts involve generating a local oscillation signal without a phase mismatch. However, according to conventional methods, implementation of a local oscillator becomes difficult or the cost of implementation increases, resulting in a limitation to removing the mismatch.  
         SUMMARY OF THE INVENTION  
         [0010]    The present invention provides a direct conversion receiver (DCR) that minimizes distortion of a converted radio frequency RF signal, by estimating a phase mismatch and gain mismatch and then calibrating for the mismatch.  
           [0011]    According to an aspect of the present invention, there is provided a direct conversion receiver comprising a transformer that converts a radio frequency (RF) signal into a differential signal; a poly-phase filter that receives the differential signal and generates an in-phase differential signal and a quadrature-phase differential signal; an in-phase mixer that mixes the in-phase differential signal with a first local oscillation signal and a second local oscillation signal, respectively, and low pass filters signals that result from mixing; a quadrature-phase mixer that mixes the quadrature-phase differential signal with the first local oscillation signal and a third local oscillation signal, respectively, and low pass filters signals that result from mixing; and a mismatch estimation unit that estimates the phase mismatch of the poly-phase filter and/or a phase mismatch of the in-phase mixer and the quadrature-phase mixer, from output signals of the in-phase mixer unit and the quadrature-phase mixer unit, so that in response to an output signal of the mismatch estimation unit, the distortion resulting from the phase mismatch of the DCR can be minimized.  
           [0012]    In the direct conversion receiver, the phase mismatch of the in-phase mixer unit and the quadrature-phase mixer unit can be adjusted in response to the output signal of the mismatch estimation unit to calibrate for mismatch in signals from the poly-phase filter. The in-phase mixer unit comprises a first mixer, which mixes the in-phase differential signal with the first local oscillation signal, and a second mixer, which mixes the in-phase differential signal with the second local oscillation signal; and the quadrature-phase mixer unit comprises a third mixer, which mixes the quadrature-phase differential signal with the first local oscillation signal, and a fourth mixer, which mixes the quadrature-phase differential signal with the third local oscillation signal. The phase difference of the second local oscillation signal and the first local oscillation signal is 90° plus a phase calibration factor, and the phase difference of the third local oscillation signal and the first local oscillation signal is 90° minus the same phase calibration factor. The in-phase mixer can be implemented using a fixed phase shifter, that phase shifts the first local oscillation signal and outputs an output signal plus a first variable phase shifter, which shifts the output signal of the fixed phase shifter by a the phase calibration factor and outputs the result as the second local oscillation signal. Similarly, the quadrature-phase mixer can be implemented using a second variable phase shifter, which shifts the output signal of the fixed phase shifter by the negative of the variable phase calibration factor and outputs the result as the third local oscillation signal.  
           [0013]    According to another aspect of the present invention, there is provided a direct conversion receiver comprising a transformer, which converts a radio frequency (RF) signal into a differential signal; a poly-phase filter that receives the differential signal, and generates an in-phase differential signal and a quadrature-phase differential signal; an in-phase mixer, which mixes the in-phase differential signal with a first local oscillation signal and a second local oscillation signal, respectively, and low pass filters the signals; a quadrature-phase mixer, which mixes the quadrature-phase differential signal with the first local oscillation signal and a second oscillation signal, respectively, and low pass filters the signals; and a mismatch estimation unit, that estimates the gain mismatch of the poly-phase filter, and/or the gain mismatch of the in-phase mixer and the quadrature-phase mixer, from the output signals of the in-phase mixer and the quadrature-phase mixer, whereby in response to the output signal of the mismatch estimation unit, the gain mismatch of the DCR can be adjusted to minimize distortion. The gain mismatch of the signals output from the in-phase mixer and the quadrature-phase mixer is adjusted relative to the other in response to the output signal of the mismatch estimation unit.  
           [0014]    According to still another aspect of the present invention, there is provided a direct conversion receiver comprising a transformer, that converts an RF signal into a differential signal; a poly-phase filter, that receives the differential signal, and generates an in-phase differential signal and a quadrature-phase differential signal; a differential signal adder which adds the in-phase differential signal and the quadrature-phase differential signal, and generates an added differential signal; a differential signal subtracter which subtracts the quadrature-phase differential signal from the in-phase differential signal, and generates a subtracted differential signal; a mixer unit that mixes the added differential signal with a first local oscillation signal and mixes the subtracted differential signal with a second local oscillation signal; and a mismatch estimation unit that estimates an entire gain mismatch of the DCR, including a gain mismatch of the poly-phase filter and a gain mismatch of the mixer unit, from the output signals of the mixer unit, so that the gain mismatch of the DCR can be minimized in response to an output signal of the mismatch estimation unit. Within the mixer unit, a first mixer that mixes the added differential signal with the first local oscillation signal and a second mixer which mixes the subtracted differential signal with the second local oscillation signal.  
           [0015]    According to yet still another aspect of the present invention, there is provided a direct conversion receiver comprising a transformer, that converts an RF signal into a differential signal; a poly-phase filter, that receives the differential signal, and generates an in-phase differential signal and a quadrature-phase differential signal; a differential signal adder, which adds the in-phase differential signal and the quadrature-phase differential signal, and generates an added differential signal; a differential signal subtracter, which subtracts the quadrature-phase differential signal from the in-phase differential signal, and generates a subtracted differential signal; a mixer unit, that mixes the added differential signal with a first local oscillation signal and mixes the subtracted differential signal with a second local oscillation signal; and a mismatch estimation unit, that estimates the gain mismatch of the poly-phase filter or the gain mismatch of the mixer unit from the output signals of the mixer unit, wherein the gain mismatch of the poly-phase filter or the mixer unit is adjusted in response to the output signal of the mismatch estimation unit.  
           [0016]    According to yet still another aspect of the present invention, component-reduced calibrated direct conversion receivers (DCRs) that minimize distortion of a converted radio frequency RF signal because they are calibrated for the phase/gain mismatch in the included mass-produced poly-phase filter but which do not contain nor need not contain a mismatch estimation unit can be mass produced for commercial purposes. These calibrated component-reduced DCRs may contain fixed or adjustable phase-shifters and/or fixed or adjustable gain adjusters for initial calibration or recalibration purposes.  
           [0017]    A method for producing such component-reduced calibrated direct conversion receivers (DCRs) comprises: providing a poly-phase filter for generating an in-phase differential signal and a quadrature-phase differential signal, having a phase mismatch and/or a gain mismatch; providing a first mixer which mixes the in-phase differential signal with a first local oscillation signal; providing a second mixer which mixes the quadrature-phase differential signal with a second local oscillation signal; and providing at least one of: a phase shifter, that phase shifts the first local oscillation signal by 90° plus the value of a phase calibration factor that calibrates for the phase mismatch of the poly-phase filter, and outputs the result as the second local oscillation signal; and/or a variable gain adjuster for adjusting the gain of the signal output from the first mixer and/or from the second mixer for calibrating for the gain mismatch of the poly-phase filter.  
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0018]    The above objects and advantages of the present invention will become more apparent by describing in detail preferred embodiments thereof with reference to the attached drawings in which:  
         [0019]    [0019]FIG. 1 is a block diagram showing a direct conversion receiver (DCR) according to a preferred embodiment of the present invention;  
         [0020]    [0020]FIG. 2 is a diagram for explaining a process of estimating and calibrating for a phase mismatch in a DCR according to a preferred embodiment of the present invention shown in FIG. 1;  
         [0021]    [0021]FIG. 3 is a diagram for explaining a process of estimating and calibrating for a gain mismatch in a DCR according to a preferred embodiment of the present invention shown in FIG. 1;  
         [0022]    [0022]FIG. 4 is a detailed diagram of a mismatch estimation unit shown in FIG. 1;  
         [0023]    [0023]FIG. 5 is a block diagram of the structure of a DCR, according to another preferred embodiment of the present invention, when a phase and gain mismatch is estimated;  
         [0024]    [0024]FIG. 6 is a block diagram of the structure of a DCR, according to another preferred embodiment of the present invention, when an RF signal is received; and  
         [0025]    [0025]FIG. 7 is a diagram showing a conventional DCR. 
     
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0026]    Referring to FIG. 1, the DCR according to a preferred embodiment of the present invention comprises a low noise amplifier  110 , a transformer  120 , a poly-phase filter  130 , a mixer unit  140 , a subtracter  181 , an adder  182 , and a mismatch estimation unit  190 .  
         [0027]    The low noise amplifier  110  receives and amplifies an RF signal. The transformer  120  converts the amplified RF signal into differential signals of 0° and 180°.  
         [0028]    The poly-phase filter  130  receives the differential signals of 0° and 180° as inputs and outputs signals having phases of 0°, 90°, 180°, and 270°. The signals of 0° and 180°, respectively, are input into the 90° and 270° inputs of the poly-phase filter  130 . The poly-phase filter  130  receives differential signals of 0° and 180° and outputs an in-phase (0° and 180°) differential signal (I_W RF ), and a quadrature-phase ( 90 ° and 270°) differential signal (Q_W RF ).  
         [0029]    The mixer unit  140  comprises in-phase first and second mixers  141  and  142 , quadrature-phase third and fourth mixers  143  and  144 , a local oscillator  151 , a 90° phase shifter  152 , variable phase shifters  153  and  154 , variable gain adjusters  161  and  162 , and low pass filters  171  through  174 . In the present embodiment, for convenience of explanation, it is assumed that the local oscillator  151 , the 90° phase shifter  152 , variable phase shifters  153  and  154 , variable gain adjusters  161  and  162 , and low pass filters  171  through  174  are included in the mixer unit  140 .  
         [0030]    The first and second in-phase mixers  141  and  142  receive the in-phase differential signal (I_W RF ) output from the poly-phase filter  130 , and mix with a first local oscillation signal (OS 1 ) and a second local oscillation signal (OS 2 ), respectively.  
         [0031]    The first and second local oscillation signals (OS 1 , OS 2 ) are oscillation signal having a predetermined oscillation frequency (W LO ). If no phase mismatch exists in the poly-phase filter  130 , it is preferable that the first and second oscillation signals (OS 1 , OS 2 ) have a 90° phase difference. However, since a phase mismatch (φ ε1 ) occurs in the poly-phase filter  130 , the phase difference of the first and second local oscillation signals (OS 1 , OS 2 ) is set to 90+φ ε2  in the present embodiment to calibrate for the phase mismatch (φ ε1 ) in the poly-phase filter  130 . Here, φ ε2  is a variable value.  
         [0032]    The local oscillator  151  generates the first local oscillation signal (OS 1 ) having the oscillation frequency (W LO ). The second local oscillation signal (OS 2 ) has a phase difference of 90+φ ε2 ° from the first local oscillation signal (OS 1 ). To generate the second local oscillation signal (OS 2 ) as described above, the first local oscillation signal (OS 1 ) is phase shifted by 90° by the 90° phase shifter  152  and then phase shifted again by φ ε2 ° by the variable phase shifter  153 . Here, φ ε2  is varied according to the output of the mismatch estimation unit  190 .  
         [0033]    Accordingly, the first mixer  141  mixes the in-phase differential signal (I_W RF ) with the first local oscillation signal (OS 1 ) and the second mixer  142  mixes the in-phase differential signal (I_W RF ) with the second local oscillation signal (OS 2 ).  
         [0034]    The third and fourth quadrature-phase mixers  143  and  144  receive the quadrature-phase differential signal (Q_W RF ) output from the poly-phase filter  130 , and mix with the first local oscillation signal (OS 1 ) and the third local oscillation signal (OS 3 ), respectively. More specifically, the third mixer  143  mixes the quadrature-phase differential signal (Q_W RF ) with the first local oscillation signal (OS 1 ) and the fourth mixer  144  mixes the quadrature-phase differential signal (Q_W RF ) with the third local oscillation signal (OS 3 ). Here, the third local oscillation signal (OS 3 ) has a phase difference of 90−φ ε2 ° from the first local oscillation signal (OS 1 ). To generate the third local oscillation signal (OS 3 ) as described above, the first local oscillation signal (OS 1 ) is phase shifted by 90° by the 90° phase shifter  152  and then phase shifted again by φ ε2 ° by the variable phase shifter  154 .  
         [0035]    As will be explained later, φ ε2  is used to calibrate for the phase mismatch (φ ε1 ) from the poly-phase filter  130 , and the value φ ε2  is varied in response to the result of phase mismatch estimation by the mismatch estimation unit  190 .  
         [0036]    The output signals of the first and second mixers  141  and  142  are sent through the first and second low pass filters  171  and  172 , respectively, to remove high frequency noise. The output signals of the third and fourth mixers  143  and  144  are sent through the variable gain adjusters  161  and  162 , respectively, and then, to remove any high frequency noise, are sent through the third and fourth low pass filters  173  and  174 . The variable gain adjusters  161  and  162  are used to calibrate for the gain mismatch from the poly-phase filter  130 . The gain is adjusted in response to the output of the mismatch estimation unit  190 . This will be explained later in detail.  
         [0037]    The subtracter  181  subtracts the output signal of the fourth low pass filter  174  from the output signal of the first low pass filter  171 , and outputs an I-path signal (I_PATH). The adder  182  adds the output signal of the second low pass filter  172  and the output signal of the third low pass filter  173 , and outputs a Q-path signal (Q_PATH). The I-path signal (I_PATH) and Q-path signal (Q_PATH) are baseband signals converted from the RF signal.  
         [0038]    The mismatch estimation unit  190  receives the I-path signal (I_PATH) and Q-path signal (Q_PATH), and estimates a phase mismatch and gain mismatch.  
         [0039]    [0039]FIG. 2 is a diagram for explaining a process of estimating and calibrating for a phase mismatch in a DCR according to a preferred embodiment of the present invention shown in FIG. 1.  
         [0040]    First, it is assumed that due to a phase mismatch (φ ε1 ) in the poly-phase filter  130  a phase difference of (φ ε1 ) occurs between differential signals (I_W RF , Q_W RF ) output from the poly-phase filter  130  occurs. Of course the phase mismatch (φ ε1 ) of the poly-phase filter  130  cannot be identified before estimation by the mismatch estimation unit  190 . To minimize the entire phase mismatch of the DCR and calibrate for the phase mismatch (φ ε1 ) of the poly-phase filter  130 , the phase mismatch (±φ ε2 ) of the mixer unit  140  is adjusted. In other words, though the degree of the phase mismatch (φ ε1 ) occurring in the poly-phase filter  130  cannot be identified, the present invention provides a method and apparatus for minimizing the entire phase mismatch of the DCR by estimating and calibrating for the phase mismatch (φ ε1 ) of the poly-phase filter  130 .  
         [0041]    For this purpose, the phase difference of the local oscillation signal (OS 1 ) provided to the first mixer  141  and the local oscillation signal (OS 2 ) provided to the second mixer  142  is set to 90+φ ε2 , where φ ε2  varies according to the output of the mismatch estimation unit  190 . Also, the phase difference of the local oscillation signal (OS 1 ) provided to the third mixer  143  and the local oscillation signal (OS 3 ) provided to the fourth mixer  144  is set to 90−φ ε2 .  
         [0042]    Alternatively, the phase difference of the local oscillation signal (OS 1 ) provided to the first mixer  141  and the local oscillation signal (OS 2 ) provided to the second mixer  142  is set to 90−φ ε2  and the phase difference of the local oscillation signal (OS 1 ) provided to the third mixer  143  and the local oscillation signal (OS 3 ) provided to the fourth mixer  144  is set to 90+φ ε2 . That is, the phase mismatch of the first mixer  141  and the second mixer  142  and the phase mismatch of the third mixer  143  and the fourth mixer  144  have the same value but with opposite signs.  
         [0043]    When the output of the mismatch estimation unit  190  is minimized, the phase mismatch (φ ε2 ) between mixers (between  141  and  142 , and between  143  and  144 ) is a value for calibrating for the phase mismatch (φ ε1 ) of the poly-phase filter  130 . Varying the phase mismatch (φ ε2 ) (also called the “phase calibration factor (φ ε2 )” between mixers (between  141  and  142 , and between  143  and  144 ) enables the search for a value at which the output of the mismatch estimation unit  190  is minimized (i.e., calibrated for phase mismatch (φ ε1 )).  
         [0044]    In the present embodiment of the invention, by adjusting the phase calibration factor (φ ε2 ) in the mixer unit  140  with respect to the estimation result of the mismatch estimation unit  190 , the entire phase mismatch of the DCR is minimized or removed. Thus, by regarding the phase mismatch (φ ε1 ) occurring in the poly-phase filter  130  as a fixed value, and by varying the phase calibration factor (φ ε2 ) in the mixer unit  140 , a value of φ ε2  is estimated at which the entire mismatch of the DCR is minimized.  
         [0045]    Alternatively, in other embodiments of the present invention, it is also possible to fix the phase mismatch (φ ε2 ) occurring in the mixer unit  140  and vary the phase mismatch (φ ε1 ) occurring in the poly-phase filter  130  such that the entire phase mismatch of the DCR can be minimized.  
         [0046]    [0046]FIG. 3 is a diagram for explaining a process of estimating and calibrating for a gain mismatch in a DCR according to a preferred embodiment of the present invention shown in FIG. 1.  
         [0047]    It is assumed that due to a gain mismatch (ΔA 1 ) in the poly-phase filter  130  a gain difference of ΔA 1  occurs between differential signals (I_W RF , Q_W RF ) output from the poly-phase filter  130  occurs. Here, it is assumed that the gain of the in-phase differential signal (I_W RF ) is greater than the gain of the quadrature-phase differential signal (Q_W RF ) by ΔA 1 . Of course the gain mismatch (ΔA 1 ) of the poly-phase filter  130  cannot be identified before estimation by the mismatch estimation unit  190 .  
         [0048]    Varying the gain mismatch (ΔA 2 ) (also called the gain calibration factor (ΔA 2 )) of the mixer unit  140  enables the discovery of a value of the gain calibration factor (ΔA 2 ) of the mixer unit  140  at which the output of the mismatch estimation unit  190  is minimized. By doing so, the gain mismatch (ΔA 1 ) of the poly-phase filter  130  is also estimated. In other words, though the degree of the gain mismatch (ΔA 1 ) occurring in the poly-phase filter  130  cannot be identified, the present invention provides a method and apparatus for minimizing the entire gain mismatch of the DCR by estimating and calibrating for the gain mismatch (ΔA 1 ) of the poly-phase filter  130 .  
         [0049]    For this purpose, variable gain adjusters  161  and  162  are disposed at the output ends of the third and fourth mixers  143  and  144 , so that the gains of the output signals of the third and fourth mixers  143  and  144  are greater than the gains of the first and second mixers  141  and  142  by ΔA 2 . The variable gain adjusters  161  and  162  can be implemented with a variable amplifier and/or a variable attenuator. Though the variable gain adjusters  161  and  162  are disposed at the output ends of the third and fourth mixers  143  and  144  in the present embodiment, in alternative embodiment of the invention they could instead be disposed at the input ends of the third and fourth mixers  143  and  144  (with or without merger), or at the input ends or output ends of the first and second mixers  141  and  142 . Thus, the variable gain adjuster or adjusters  161  and  162  will be disposed on the path of the in-phase differential signal (I_W RF ) and/or the path of the quadrature-phase differential signal (Q_W RF ) so that the gain mismatch between the path of the in-phase differential signal (I_W RF ) and the path of the quadrature-phase differential signal (Q_W RF ) is ΔA 2 . The “path of the in-phase differential signal (I_W RF )” indicates a path from the output end of the poly-phase filter  130  through the first and second mixers  141  and  142  to the adder  182  and the subtracter  181 , while the “path of the quadrature-phase differential signal (Q_W RF )” indicates a path from the output end of the poly-phase filter  130  through the third and fourth mixers  143  and  144  to the adder  182  and the subtracter  181 .  
         [0050]    Accordingly, varying the gain mismatch in the mixer unit  140 , that is, varying the gain calibration factor ΔA 2  between the path of the in-phase differential signal (I_W RF ) and the path of the quadrature-phase differential signal (Q_W RF ) enables the search for a value of ΔA 2  at which the output of the mismatch estimation unit  190  is minimized. When the output of the mismatch estimation unit  190  is minimized, the gain mismatch (ΔA 2 ) of the mixer unit  140  is a value for calibrating for the gain mismatch ΔA 1  of the poly-phase filter  130 .  
         [0051]    In the present embodiment of the invention, by adjusting the gain mismatch (ΔA 2 ) in the mixer unit  140 , with respect to the estimation result of the mismatch estimation unit  190 , the entire gain mismatch of the DCR is minimized or removed. Thus, by regarding the gain mismatch ΔA 1  occurring in the poly-phase filter  130  as a fixed value, and by varying the gain mismatch (ΔA 2 ) of the mixer unit  140 , a value of gain calibration factor (ΔA 2 ) is estimated at which the entire gain mismatch of the DCR is minimized.  
         [0052]    Alternatively, in other embodiments of the invention, it is also possible to fix the gain mismatch (ΔA 2 ) occurring in the mixer unit  140  and vary the gain mismatch (ΔA 1 ) of the poly-phase filter  130  such that the entire gain mismatch of the DCR can be minimized.  
         [0053]    In order to explain the principle by which the entire gain mismatch and phase mismatch of the DCR are minimized through the method described above, it is assumed that a cosine signal having a predetermined frequency (WRF) is input as an RF receiving signal to the DCR, according to the exemplary embodiment of the invention shown in FIG. 1. Then, signals (II, IQ, QI, QQ) output from the first through fourth low pass filters  171  through  174  can be expressed by the following equation 1:  
           II ( t )=(1+Δ A   1 ) cos(Δ wt )/4  
           IQ ( t )=(1 +ΔA   1 ) sin(Δ wt+φ   ε2 )/4  
           QI ( t )=(1 +ΔA   2 ) sin(Δ wt+φ   ε1 )/4  
           QQ ( t )=(1 +ΔA   2 ) cos(Δ wt+φ   ε1−φ   ε2 )/4  (1)  
         [0054]    As described above, ΔA 1  indicates the gain mismatch in the poly-phase filter  130  and φ ε1  indicates the phase mismatch in the poly-phase filter  130 . ΔA 2  indicates the gain mismatch in the mixer unit  140  and is used for calibrating for the gain mismatch (ΔA 1 ) in the poly-phase filter  130 . φ ε2  indicates the phase mismatch in the mixer unit  140  and is used for calibrating for the phase mismatch (φ ε1 ) in the poly-phase filter  130 .  
         [0055]    Since the I-path signal (I_PATH) can be obtained by subtracting the output signal (QQ) of the fourth low pass filter  174  from the output signal (II) of the first low pass filter  171 , the I-path signal (I_PATH) can be expressed as the following equation 2:  
                       (     II   -   QQ     )          (   t   )       =              1   4     [         (     1   +     Δ                   A   1         )          cos        (     Δ                 wt     )         -                              (     1   +     Δ                   A   2         )          cos        (       Δ                 wt     +     ϕ   ɛ1     -     ϕ   ɛ2       )         ]               =                1   4          [       (     1   +     Δ                   A   1         )     -       (     1   +     Δ                   A   2         )          cos        (       ϕ   ɛ1     -     ϕ   ɛ2       )           ]            cos        (     Δ                 wt     )                     =                1   4          [       (     1   +     Δ                   A   2         )          sin        (       ϕ   ɛ1     -     ϕ   ɛ2       )         ]            sin        (     Δ                 wt     )                       (   2   )                               
 
         [0056]    Since the Q-path signal (Q_PATH) can be obtained by adding the output signal (IQ) of the second low pass filter  172  and the output signal (QI) of the third low pass filter  173 , the Q-path signal (Q_PATH) can be expressed as the following equation 3:  
                       (     IQ   -   QI     )          (   t   )       =              1   4     [         (     1   +     Δ                   A   1         )          sin        (       Δ                 wt     +     ϕ   ɛ2       )         -                              (     1   +     Δ                   A   2         )          sin        (       Δ                 wt     +     ϕ   ɛ1       )         ]               =              1   4     [         (     1   +     Δ                   A   1         )          cos        (     ϕ   ɛ2     )         -                                (     1   +     Δ                   A   2         )          cos        (     ϕ   ɛ1     )         ]          sin        (     Δ                 wt     )                   =              1   4     [         (     1   +     Δ                   A   1         )          sin        (     ϕ   ɛ2     )         -                                (     1   +     Δ                   A   2         )          cos        (     ϕ   ɛ1     )         ]          cos        (     Δ                 wt     )                     (   3   )                               
 
         [0057]    The mismatch estimation unit  190  squares the I-path signal (II-QQ) and the Q-path signal (IQ+QI), respectively, and adds the two squared signals.  
         [0058]    [0058]FIG. 4 is a detailed diagram of the mismatch estimation unit  190  shown in FIG. 1. Referring to FIG. 4, the mismatch estimation unit  190  comprises square units  191  and  192  for squaring the I-path signal (II-QQ) and the Q-path signal (IQ+QI) respectively, an adder  193 , and a low pass filter  194 . The mismatch estimation unit  190  squares the I-path signal (II-QQ) and the Q-path signal (IQ+QI) respectively, adds the two squared signals, low pass filters the added signal, and outputs the result as an estimated mismatch value. The output of the mismatch estimation unit is fed back to the poly-phase filter  130  or the mixer unit  140  so that the phase/gain mismatch in the poly-phase filter  130  or the mixer unit  140  can be adjusted.  
         [0059]    In the following expression 4,  
         Δ A   1   ,ΔA   2 ,(Δ A   1−ΔA   2 )&lt;&lt;1, sin(φ ε1 −φ ε2 )≅φ ε1 −φ ε2 , cos(φ ε1 −φ ε2 )≅1  (4)  
         [0060]    it is assumed that the gain mismatch (ΔA 1 ) of the poly-phase filter  130 , the gain mismatch (ΔA 2 ) of the mixer unit  140 , the difference (ΔA 1 −ΔA 2 ) of the two gains, and the difference (φ ε1 −φ ε2 ) of the phase mismatch of the poly-phase filter  130  and the phase mismatch of the mixer unit  140  are much less than 1. If the conditions of expression 4 are applied to equations 2 and 3, equations 2 and 3 are expressed simply as the following equation 5:  
                   (     II   -   QQ     )          (   t   )       ≅       1   4          [         (       Δ                   A   1       -     Δ                   A   2         )          cos        (     Δ                 wt     )         +       (       ϕ   ɛ1     -     ϕ   ɛ2       )          sin        (     Δ                 wt     )           ]              
              (     IQ   +   QI     )          (   t   )       ≅       1   4          [         (       Δ                   A   1       -     Δ                   A   2         )          sin        (     Δ                 wt     )         -       (       ϕ   ɛ1     -     ϕ   ɛ2       )          cos        (     Δ                 wt     )           ]                 (   5   )                               
 
         [0061]    If equation 5 is used, the value obtained by squaring the I-path signal and the Q-path signal, respectively, is expressed as the following equation 6:  
                   (     II   -   QQ     )     2     +       (     IQ   +   QI     )     2       =       1   16          [         (       Δ                   A   1       -     Δ                   A   2         )     2     +       (       ϕ   ɛ1     -     ϕ   ɛ2       )     2       ]               (   6   )                               
 
         [0062]    Equation 6 shows that there is only one pair of estimated variables (i.e., correction factors) that can minimize the value obtained by squaring the I-path signal and the Q-path signal, respectively, and adding the results. Each of the variables can be obtained independently of each other. Thus, in equation 6, when each of the squared terms is minimized, the entire value is minimized. Also, since the gain variable and the phase variable are independent of each other, each variable can be obtained by fixing one squared term and varying the other squared term. Accordingly, the amount of computation decreases greatly.  
         [0063]    [0063]FIG. 5 is a block diagram of a DCR according to another preferred embodiment of the present invention. The DCR shown in FIG. 5 has a structure for estimating and compensating for the phase and gain mismatch, that is, a structure for calibrating for the phase and gain mismatch.  
         [0064]    Referring to FIG. 5, the DCR according to the embodiment of the present invention comprises a low noise amplifier  110 , a transformer  120 , a poly-phase filter  130 , an adder  511 , a subtracter  512 , a mixer unit  520 , and a mismatch estimation unit  550 .  
         [0065]    Since the functionality of the low noise amplifier  110 , the transformer  120 , and the poly-phase filter  130  are the same as explained in reference to FIG. 1, a detailed explanation will be omitted here.  
         [0066]    The adder  511  adds the in-phase differential signal (I_W RF ) and the quadrature-phase differential signal (Q_W RF ) output from the poly-phase filter  130  and outputs the result (AS). The subtracter  512  subtracts the quadrature-phase differential signal (Q_W RF ) from the in-phase differential signal (I_W RF ) and outputs the result (SS).  
         [0067]    The mixer unit  520  comprises a first mixer  521 , a second mixer  522 , a local oscillator  531 , a 90° phase shifter  532 , a variable phase shifter  533 , a variable gain adjuster  523 , and low pass filters  541  and  542 . For convenience of explanation, in the present embodiment, it is assumed that the local oscillator  531 , the 90° phase shifter  532 , the variable phase shifter  533 , the variable gain adjuster  523 , and low pass filters  541  and  542  are included in the mixer unit  520 .  
         [0068]    The first mixer  521  receives a signal (AS), obtained by adding the in-phase differential signal (I_W RF ) and the quadrature-phase differential signal (Q_W RF ), and mixes this signal (AS) with a first local oscillation signal (OS 1 ). The second mixer  522  receives a signal (SS), obtained by subtracting the quadrature-phase differential signal (Q_W RF ) from the in-phase differential signal (I_W RF ), and mixes this signal (SS) with a second local oscillation signal (OS 2 ).  
         [0069]    The first and second local oscillation signals (OS 1 , OS 2 ) have a predetermined oscillation frequency (W LO ). The local oscillator  531  generates the first local oscillation signal (OS 1 ) having the oscillation frequency (W LO ). The second local oscillation signal (OS 2 ) has a phase difference of 90+φ ε2 ° from the first local oscillation signal (OS 1 ). In order to generate the second local oscillation signal (OS 2 ) as described above, the first local oscillation signal (OS 1 ) is phase shifted by 90° by the 90° phase shifter  532  and then phase shifted again by φ ε2 ° by the variable phase shifter  533 . φ ε2  is used to calibrate for the phase mismatch (φ ε1 ) in the poly-phase filter  130 , and the value φ ε2  varies in response to the output of the mismatch estimation unit  190 .  
         [0070]    To remove high frequency band noise, the output signal of the first mixer  521  is sent through the first low pass filter  541 . The output signal of the second mixer  522  is sent through the variable gain adjuster  523 , and then to remove high frequency band noise, is sent through the second low pass filter  542 . The variable gain adjuster  523  is used to calibrate for the gain mismatch in the poly-phase filter  130  and adjusts the gain in response to the output of the mismatch estimation unit  550 .  
         [0071]    The mismatch estimation unit  550  receives the output signals (I_PATH′, Q_PATH′) of the first and second low pass filters  541  and  542  and estimates the phase mismatch and gain mismatch.  
         [0072]    The mismatch estimation unit  550  comprises adders  551  and  552 , third and fourth mixers  561  and  562 , square units  571  and  572 , and a low pass filter  553 . The adder  551  adds the output signals (I_PATH′, Q_PATH′) of the first and second low pass filters  541  and  542 .  
         [0073]    The third and fourth mixers  561  and  562  mix the output signal of the adder  551  with predetermined local oscillation signals (OS 4 , OS 5 ), respectively. Here, the local oscillation signal (OS 4 ) input to the third mixer  561  and the local oscillation signal (OS 5 ) input to the fourth mixer  562  have an identical oscillation frequency and a 90° phase difference. The third and fourth mixers  561  and  562  induce a 90° phase difference in the paths of the output signal of the adder  551 . Accordingly, instead of the third and fourth mixers  561  and  562 , a phase shifter can be used in the path from mixer  551  to one of square units  571  and  572 ., In such alternative embodiments of the invention, the output signal of the adder  551  is directly input to one square unit  571 , and also is phase shifted by 90° and then input to the other square unit  572 .  
         [0074]    The square units  571  and  572  square each of pair of phase-shifted signals derived from adder  551  (e.g., square the output signals of the third mixer  561  and the fourth mixer  562 , respectively). The adder  552  adds the output signals of the square units  571  and  572 . The output signal of the adder  552  is sent through the low pass filter  553  and output as an estimated mismatch value. The estimated mismatch value is fed back to the poly-phase filter  130  or the mixer unit  520  so that the phase/gain mismatch in the poly-phase filter  130  or the mixer unit  520  can be adjusted or calibrated for in the mixer unit  520 .  
         [0075]    By constructing the DCR as shown in FIG. 5, the output of the mismatch estimation unit  550  is the same as in equation 6 (above). Accordingly, the process for estimating gain and phase calibration factors that minimize the output of the mismatch estimation unit  550  is similar to the process employed in the first embodiment of the present invention (shown in FIG. 1).  
         [0076]    The process for estimating and calibrating for the phase mismatch in a DCR, according to another embodiment of the present invention shown in FIG. 5, will now be explained.  
         [0077]    It is assumed that, due to a phase mismatch (φ ε1 ) in the poly-phase filter  130 , a phase difference of φ ε1  between differential signals (I_W RF , Q_W RF ) output from the poly-phase filter  130  occurs. The phase mismatch (φ ε1 ) of the poly-phase filter  130  is a fixed value. To calibrate for this, and in order to minimize the entire phase mismatch of the DCR, the phase calibration factor (φ ε2 ) of the mixer unit  520  is adjusted.  
         [0078]    For this purpose, the phase difference of the local oscillation signal (OS 1 ) provided to the first mixer  521  and the local oscillation signal (OS 2 ) provided to the second mixer  522  is set to 90+φ ε2 , and φ ε2  varies according to the result of the mismatch estimation unit  550 .  
         [0079]    Varying the phase mismatch (φ ε2 ) between mixers  521  and  522  enables the search for a value at which the output of the mismatch estimation unit  550  is minimized. When the output of the mismatch estimation unit  550  is minimized the phase mismatch (φ ε2 ) between mixers  521  and  522  is a value for calibrating for the phase mismatch (φ ε1 ) of the poly-phase filter  130 .  
         [0080]    Also in the embodiment of the present invention, by adjusting the phase calibration factor (φ ε2 ) in the mixer unit  520  with respect to the estimation result of the mismatch estimation unit  550 , the entire phase mismatch of the DCR is minimized or removed. Thus, by regarding the phase mismatch (φ ε1 ) occurring in the poly-phase filter  130  as a fixed value, and by varying the phase calibration factor (φ ε2 ) in the mixer unit  520 , a value is estimated at which the entire mismatch of the DCR is minimized.  
         [0081]    Alternatively, in other embodiments of the invention, it is also possible to fix the phase mismatch (φ ε2 ) occurring in the mixer unit  520  and vary the phase mismatch (φ ε1 ) occurring in the poly-phase filter  130  such that the entire phase mismatch of the DCR can be minimized.  
         [0082]    The process for estimating and calibrating for the gain mismatch in the DCR according to the embodiment of the present invention shown in FIG. 5, will now be explained.  
         [0083]    It is assumed that, due to a gain mismatch ΔA 1  in the poly-phase filter  130 , a gain difference of ΔA 1  between differential signals (I_W RF , Q_W RF ) output from the poly-phase filter  130  occurs. Here, it is assumed that the gain of the in-phase differential signal (I_W RF ) is greater than the gain of the quadrature-phase differential signal (Q_W RF ) by ΔA 1 . Of course the gain mismatch (ΔA 1 ) of the poly-phase filter  130  cannot be identified before estimation by the mismatch estimation unit  550 .  
         [0084]    By varying the gain calibration factor (ΔA 2 ) of the mixer unit  520 , a value of the gain mismatch (ΔA 2 ) of the mixer unit  520  is found at which the output of the mismatch estimation unit  550  is minimized and by doing so, an estimate is achieved for calibrating for the gain mismatch (ΔA 1 ) of the poly-phase filter  130 .  
         [0085]    The variable gain adjuster  523  is disposed at the output end of the second mixer  522  so that the gain of the output signal of the second mixer  522  is greater than the gain of the output signal of the first mixer  521  by ΔA 2 . The variable gain adjuster  523  can be implemented with a variable amplifier and/or a variable attenuator. Though the variable gain adjuster  523  is disposed at the output end of the second mixer  522  in the present embodiment, it can also be disposed at other places as explained in the description of the first exemplary embodiment of the present invention.  
         [0086]    Accordingly, by varying the gain calibration factor ΔA 2  in the mixer unit  520  enables the search for a value at which the output of the mismatch estimation unit  550  is minimized. The value of the gain calibration factor (ΔA 2 ) of the mixer unit  520  when the output of the mismatch estimation unit  550  is minimized is a value for calibrating for the gain mismatch ΔA 1  of the poly-phase filter  130 .  
         [0087]    In the present embodiment of the invention, by adjusting the gain calibration factor (ΔA 2 ) in the mixer unit  520  with respect to the estimation result of the mismatch estimation unit  550 , the entire gain mismatch of the DCR is minimized or removed. Thus, by regarding the gain mismatch (ΔA 1 ) occurring in the poly-phase filter  130  as a fixed value, and by varying the gain calibration factor ΔA 2  of the mixer unit  520 , a value for ΔA 2  is estimated at which the entire gain mismatch of the DCR is minimized.  
         [0088]    Alternatively, in other embodiments of the invention, it is also possible to fix the gain mismatch (ΔA 2 ) occurring in the mixer unit  520  and vary the gain mismatch ΔA 1  of the poly-phase filter  130  such that the entire gain mismatch of the DCR is minimized.  
         [0089]    The structure of the DCR shown in FIG. 5 is provided for estimating and calibrating for the entire gain mismatch and phase mismatch of the DCR. After estimating and calibrating for the gain mismatch and phase mismatch by using the DCR structure shown in FIG. 5, minimally distorted baseband signals (I_PATH, Q_PATH) that have been directly converted into baseband signals from the RF signal may be obtained through the simplified (and calibrated) DCR structure shown in FIG. 6.  
         [0090]    The DCR of FIG. 6, does not include the mismatch estimation unit  550 , the adder  511 , and the subtracter  512 , that were provided in the DCR of FIG. 5 used for estimating and calibrating for the phase/gain mismatch of the poly-phase filter  130 . Here, the second local oscillation signal (OS 2 ) has the same oscillation frequency as that of the first local oscillation signal (OS 1 ). The phase difference between the first and second local oscillation signals (OS 1 , OS 2 ) is set to 90 degrees plus φ ε2  (e.g., where φ ε2  is the predetermined value of the calibration factor determined appropriate for the particular poly-phase filter  130  in accordance with the methods of the present invention) to calibrate for the phase mismatch (φ ε1 ) in signals from the poly-phase filter  130 . Here, φ ε2  may be a fixed or a variable value. Similarly, the gain mismatch (ΔA 2 ) may be a fixed or a variable value to calibrate for the gain mismatch (ΔA 1 ) in the poly-phase filter  130  (as may be predetermined for the particular poly-phase filter  130  in accordance with the methods of the present invention). Accordingly, the circuits (e.g., gain adjusters and phase-shifters) for implementing calibration values φ ε2  and ΔA 2  respectively may be fixed or variable gain/phase circuits.  
         [0091]    Accordingly, the in-phase differential signal (I_W RF ) output from the poly-phase filter  130  is output as the I-path signal (I_PATH) through the first mixer  521  and the first low pass filter  541 . The quadrature-phase differential signal (Q_W RF ) output from the poly-phase filter  130  is output as the Q-path signal (Q_PATH) through the second mixer  522 , the variable gain adjuster  523 , and the second low pass filter  542 .  
         [0092]    According to the present invention, the phase mismatch and gain mismatch of a DCR is removed or minimized. Accordingly, distortion of the radio frequency signal (RF) that is directly converted by the DCR of the present invention is minimized.  
         [0093]    Exemplary embodiments of the inventive method and apparatus have been described and explained above. However, the present invention is not limited to the preferred embodiments described above, and it is apparent that variations and modifications by those skilled in the art can be effected within the spirit and scope of the present invention defined in the appended claims. Therefore, the scope of the present invention is not determined by the above description but by the accompanying claims.