Abstract:
The present invention is a circuit and method for reducing switching and reverse recovery losses in the output rectifiers while creating zero voltage switching conditions for the primary switchers. There are described two output configurations, one employing a soft commutation inductor element a bridge rectifier and a output filter capacitor, the second using a soft commutation inductor element a rectification-filtering bridge composed by tow capacitors and two capacitors. Both secondary circuits can be driven by three primary circuits. A first circuit is a full bridge with phase shift control, and a second circuit is a half bridge topology with an additional bydirectional switch which achieves two goals, on to get soft switching commutation across all the primary switches, the second to create the right waveforms in the secondary suitable with the claims in this invention. The third topology is a phase shifted two transistors forward. The circuits claimed in this invention can provide soft commutation across the primary switching elements and secondary rectifier means, clamping the voltage across the rectifiers to the output voltage eliminating the need for snubbers circuits both in primary and the secondary section.

Description:
BACKGROUND OF INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    This invention relates to DC-to-DC converters, DC-to-AC inverters and AC-to-DC converters. The major characteristic of this power conversion technique is that primary switching elements switches at zero voltage and the secondary rectifiers means have negligible reverse recovery losses.  
           [0003]    2. Description of the Prior Art  
           [0004]    There is a continuing industry demand for increasing power density, which means more power transferred in a given volume. A method for increasing the power transfer through the converter is to increase the switching frequency in order to minimize the size of magnetic and the capacitors. Using prior art topologies such as forward or flyback, which employ “hard” switching techniques makes high frequency operation less efficient. The switching losses associated with switching elements, which turn on when there is a voltage across them, are proportional with the switching frequency. An increase in switching frequency leads to an increase in switching losses and an increase in level of electromagnetic interference (EMI).  
           [0005]    In order to overcome limitations in switching speeds, the prior art has devised a new family of soft transition. The U.S. Pat. Nos. 5,132,889 , 5,126,931 , 5,231,563 , 5,434,768 present several methods of accomplishing zero voltage switching across the primary switches.  
           [0006]    Another power loss mechanism is due to the reverse recovery in the output rectifiers. During switching when a negative polarity voltage is applied to a rectifier in conduction the current through the rectifier will continue to conduct until all the carriers in the rectifier&#39;s junctions are depleted. During this period of time the current polarity will reverse, the current flowing form the cathode to the anode, while the voltage across the diode is still positive from the anode to the cathode. The current flowing in reverse through the diode will reach a peak value referred in literature as Irrm. Further on, while the rectifiers&#39; junction is depleting the carriers, the rectifier becomes a high impedance device. The current through the rectifier will decrease rapidly from lrrm level to zero. During the same time the negative voltage across the rectifier will build up to high levels.  
           [0007]    During the period of time when there is a negative voltage across the diode and negative current is flowing through it, there will be power dissipation in the device. This kind of loss is referred in the literature as reverse recovery losses. The reverse recovery loss is proportional with the reverse recovery current Irrm, the negative voltage across the rectifier and the frequency.  
           [0008]    The reverse recovery current Irrm, which is a key component in reverse recovery loss, is function of the type of device, the temperature and the current slope at turn off. The reverse recovery characteristics are getting worst for higher voltage rectifiers. As a result the reverse recovery loss becomes a significant loss mechanism for higher output voltage applications. The reverse recovery current lrrm is direct dependent of the current slope at turn off. A soft slope reduces the reverse recovery current and as a consequence reduces the reverse recovery loss. To accomplish a very soft slope current at turn off an inductive element has to be in series with the rectifier. The inductor element will prevent a fast current variation dI/dt. The presence of an inductive element in series with the rectifier will increase the negative voltage across the rectifier at turn off. The reverse voltage across the rectifier can reach very high levels and can exceed the voltage break down of the device, leading to failure.  
           [0009]    RC snubbers or complicated lossless snubers can be added across the rectifier to reduce the reverse recovery loss and the voltage stress on the devices. This leads to complex circuits and which negatively affects the efficiency and the reliability. As a result of these limitations the high voltage converters have to operate at lower frequency in order to reduce the power dissipation associated with reverse recovery.  
           [0010]    What is needed is a converter topology which can operate at constant frequency with zero voltage switching on the primary switches and soft commutation on the output rectifiers, wherein low current slope through the rectifiers at turn off is associated with low negative voltage across the rectifiers. The lowest voltage across the output rectifiers in a DC-DC converter is the output voltage. As a result our goal is to reduce the negative voltage across the output rectifier to the level of the output voltage.  
         BRIEF SUMMARY OF THE INVENTION  
         [0011]    The invention applies to topologies in which the voltage in the secondary of the transformer has three states. One state wherein the voltage in the secondary is a positive voltage source, another stage wherein the voltage in the secondary of the transformer is zero with a very low internal impedance and the third stage wherein the voltage in the secondary of the transformer is a negative voltage source. To provide such secondary signal we have identified three topological structures. One is the full bridge phase shifted topology. The second topology is a half bridge utilizing and additional bydirectional switch, depicted in FIG. 9A. The third topology is two transistors forward phase shifted, depicted in FIG. 14A.  
           [0012]    All these topologies operate in a similar manner. An input voltage source is applied to the primary of a transformer through controlled switching elements. The primary winding of the transformer has two terminations. For simplicity we are going to refer to on end of the primary winding of the transformer as a dotted end. The secondary winding of the transformer has also two terminations. When a voltage is applied to the primary winding of the transformer with the positive polarity at the dotted end, a voltage will be induced in the secondary. The termination of the secondary winding where the voltage induced has a positive polarity is referred as a dotted end of the secondary winding.  
           [0013]    The input voltage source is applied to the primary winding of the transformer through two controlled switching elements. A control-switching element is an electronic switch, which can be controlled by a control signal to exhibit low impedance when the switching element is turned ON or large impedance when the switching element is turned OFF. The input voltage source is applied to the primary winding through two controlled switching devices, which connects the termination of the transformer to the termination of the input voltage source selectively. The dotted end of the primary winding of the transformer can be connected to the positive end of the input voltage source and the other end of the primary winding of the transformer is connected to the negative end of the input voltage source. This operation will be further referred to as positive voltage across the primary winding.  
           [0014]    The controlled switching elements can connect also the dotted end of the primary winding of the transformer to the negative end of the input voltage source and the other end of the primary winding of the transformer is connected to the positive end of the input voltage source. This operation will be further referred as negative voltage across the primary winding.  
           [0015]    The control switching elements can also short out the primary winding of the transformer by applying low impedance across the winding. This operation will be further referring to as the dead time.  
           [0016]    The controlled switching elements can be controlled in a such way to apply sequentially a positive voltage across the primary winding for given period of time, referred as positive ON time, short the primary winding for a period of time, referred as dead time, apply a negative voltage across the primary winding for a given period of time, referred as negative ON time, equal as duration with the positive ON time. If the summation of positive ON time, dead time and negative ON time is constant, the mode of operation is referred as constant frequency operation.  
           [0017]    The power converter can also operate in frequency modulation mode, wherein the summation of positive ON time, dead time and negative ON time is not constant. We introduce the term of duty cycle, which is defined as the ratio between the summation of positive and negative ON time and the summation of the positive ON time, twice the dead time and negative ON time. By varying the duty cycle the power transferred through the transformer can be controlled. The duty cycle can be varied by varying the duration of the positive and negative ON time, for the constant frequency operation. For variable frequency operation the duty cycle control can be made by maintaining the negative and positive ON time constant and varying the dead time, or by varying the positive and negative ON time and maintain the dead time constant or by varying the positive and negative ON time and the dead time in the same time. Important is to have the positive ON time equal to the negative ON time. Another important element of this technology is the low bydirectional impedance across the secondary winding of the transformer, wherein the secondary current can flow freely in both directions.  
           [0018]    One key element in this invention is an additional inductor element in series with the secondary winding, labeled soft commutation inductor. The inductor can be also located in the primary section in series with the primary winding of the transformer. The soft commutation inductive element can be also split, one section located in the primary, in series with the primary winding and an another section in the secondary in series with the secondary winding. In the case when the soft commutation inductor is located in the secondary, there is a bydirectional rectification means connected in series with it and the secondary winding. A bridge of rectifiers can form the bydirectional rectification means. Across the capacitor element is connected the load. The bridge of rectifiers has a first input terminal a second input terminal a first output terminal and a second output terminal. The first rectifier is connected between the first input terminal and the first output terminal with the cathode to the first output terminal, the second rectifier is connected between the second input terminal and the first output terminal, with the cathode to the first output terminal. The third rectifier is connected between the second input terminal and the second output terminal with the cathode to the second input terminal, the fourth rectifier is connected between the first input terminal and the second output terminal with the cathode to the first input terminal. The AC voltage source in series with said inductive element is connected between the first input terminal and the second input terminal. The output capacitor is in parallel with the load is connected between the first output terminal and the second output terminal.  
           [0019]    The bydirectional rectification means can be also constructed using two rectifiers and two capacitors. The bridge of rectifiers means and capacitors having a first input terminal a second input terminal a first output terminal and a second output terminal. The first rectifier is connected between the first input terminal and the first output terminal with the cathode to the first output terminal, the second rectifier is connected between the first input terminal and the second output terminal, with the cathode to the first input terminal. The first capacitor is connected in between the first output terminal and the second input terminal, and the second capacitor is connected between the second input terminal and the second output terminal. The said AC voltage source is in series with the soft commutation inductive element and connected between the first input terminal and the second input terminal. The load is connected between the first output terminal to the second output terminal.  
           [0020]    During the positive and negative ON time the power is transferred from the primary to the secondary via the transformer, the soft commutation inductor, and the bydirectional rectifier means to the load. In the same time energy is stored in the soft commutation inductor. During the dead time, the energy stored in the soft commutation inductor is further transferred to the load. There are two modes of operation. One mode of operation referred as discontinuous conduction mode, the entire energy stored in the soft commutation inductor is transferred to the load prior the change of the voltage polarity on the transformer. The second mode of operation referred as continuous mode, there is still energy left in the soft commutation inductor prior the reversal of the voltage polarity in the transformer. The discontinuous mode of operation has the advantage of transferring the energy from the primary to the secondary unidirectional at each cycle. The continuous mode of operation will transfer the energy left in the soft commutation diode back to the primary before the next energy transfer from primary to the secondary starts.  
           [0021]    A critical conduction mode of operation can be implemented wherein the reversal of the voltage polarity in the transformer is accomplished just after the entire energy in the soft commutation inductor is transferred to the secondary. This leads to a modulation in frequency, wherein the frequency will increase at light loads, and decrease at heavy loads. A mix mode of operation can be also implemented wherein some high frequency boundary or low frequency boundaries or both are set. There are several major advantages of this topology.  
           [0022]    One of the major advantages is the fact that the voltage across the rectifiers is clamped to the output voltage. There is not ringing or spike across the rectifiers which exceed the output voltage. The voltage across the rectifiers for a given output voltage is the lowest theoretical possible. In most of the topologies operating over a range of input and output voltages the voltage across the rectifiers can be several times larger that the output voltage. For a single ended forward converter the output voltage is Vr=(Vin Max /Vin Min )*V O /D Max , wherein Vin Max  &amp; Vin Min  is the maximum and minimum input voltage and D Max  is the maximum duty cycle, and Vo is the output voltage. For an input voltage range of 2:1 and 50% maximum duty cycle, the reverse voltage across the rectifier is 4*Vo . In conclusion in these topologies we achieve the lowest voltage across the rectifiers for a given output voltage.  
           [0023]    Another major advantage of this topology is the fact that the current slope through the rectifier at turn off is controlled by the soft commutation inductor. As a result there is a controlled dI/dt. A soft current through the rectifier at turn OFF reduces considerably the reverse recovery current. The clamped voltage across the output rectifiers in association with the soft current slope at turn OFF leads to reduced reverse recovery losses. If the circuits operate in continuous mode the reverse recovery losses are reduced, and if we operate in discontinuous conduction mode the reverse recovery losses are actually eliminate. This is due to the fact that the current through the rectifiers reaches zero prior the reverse voltage is applied to them.  
           [0024]    The invention can be better visualized by turning to the following drawings.  
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0025]    [0025]FIG. 1 is a schematic of a converter utilizing the power transfer methodology of the invention.  
         [0026]    [0026]FIG. 1B is a timing diagram of the circuit of FIG. 1.  
         [0027]    [0027]FIG. 2 is a schematic diagram of an AC-DC converter wherein prior art technique is illustrated.  
         [0028]    [0028]FIG. 3A is a schematic diagram of a DC-DC Converter wherein another prior art technique is illustrated.  
         [0029]    [0029]FIG. 3B is a timing diagram of the circuit of FIG. 3A.  
         [0030]    [0030]FIG. 4A is a schematic diagram of an AC-DC Converter using an embodiment of this invention.  
         [0031]    [0031]FIG. 4B is a timing diagram of the circuit of FIG. 4A.  
         [0032]    [0032]FIG. 5A is a schematic diagram of an AC-DC Converter using another embodiment of this invention.  
         [0033]    [0033]FIG. 5B is a timing diagram of the circuit of FIG. 5A.  
         [0034]    [0034]FIG. 6A is a schematic diagram of an AC-DC Converter using an embodiment of this invention depicted in FIG. 4A, operating in continuous mode.  
         [0035]    [0035]FIG. 5B is a timing diagram of the circuit of FIG. 6A.  
         [0036]    [0036]FIG. 7A is a schematic diagram of an AC-DC Converter using another embodiment of this invention.  
         [0037]    [0037]FIG. 7B is a timing diagram of the circuit of FIG. 7A.  
         [0038]    [0038]FIG. 8A is a schematic diagram of a DC-AC Converter, utilizing a phase shift bridge topology suitable with the AC-DC converters depicted in several embodiments of the invention.  
         [0039]    [0039]FIG. 8B is a timing diagram of the circuit of FIG. 8A.  
         [0040]    [0040]FIG. 9A is a schematic diagram of an DC-AC Converter, utilizing a half bridge topology employing an additional bydirectional switch, topology suitable with the AC-DC converters depicted in several embodiments of the invention.  
         [0041]    [0041]FIG. 9B is a timing diagram of the circuit of FIG. 9A.  
         [0042]    [0042]FIG. 10A is a schematic diagram of an AC-DC Converter depicted in FIG. 4A wherein two of the rectifiers are replaced by synchronous rectifiers.  
         [0043]    [0043]FIG. 10B is a timing diagram of the circuit of FIG. 10A.  
         [0044]    [0044]FIG. 11A is a schematic diagram of an AC-DC Converter depicted in FIG. 4A wherein all of the rectifiers are replaced by synchronous rectifiers.  
         [0045]    [0045]FIG. 11B is a timing diagram of the circuit of FIG. 11A.  
         [0046]    [0046]FIG. 12A is a schematic diagram of an AC-DC Converter depicted in FIG. 5A wherein all of the rectifiers are replaced by synchronous rectifiers.  
         [0047]    [0047]FIG. 12B is a timing diagram of the circuit of FIG. 12A.  
         [0048]    [0048]FIG. 13A is a schematic diagram of an DC-DC Converter using the embodiment presented in FIG. 4A, wherein the soft commutation inductor is transferred in primary section in series with the primary winding. The AC signal across the primary winding connected in series with the soft commutation inductor is produced by a phase shift full bridge topology depicted in FIG. 8A.  
         [0049]    [0049]FIG. 13B is a schematic diagram of an DC-DC Converter using the embodiment presented in FIG. 4A, wherein the soft commutation inductor is split in two sections, one section transferred in primary section in series with the primary winding, and the other section in the secondary. The AC signal across the primary winding connected in series with one section of the soft commutation inductor is produced by a phase shift full bridge topology, depicted in FIG. 8A. This structure being another embodiment of the invention.  
         [0050]    [0050]FIG. 13C is a schematic diagram of an DC-DC Converter using the embodiment presented in FIG. 4A, wherein the soft commutation inductor is transferred in primary section in series with the primary winding and providing a center tap, connected to a capacitor. The AC signal across the primary winding connected in series with one section of the soft commutation inductor is produced by a phase shift full bridge topology, depicted in FIG. 8A. This structure being another embodiment of the invention.  
         [0051]    [0051]FIG. 13D is a schematic diagram of an DC-DC Converter using the embodiment presented in FIG. 4A, wherein the transformer is implemented by using two identical transformers connected in series. The additional circuit formed by an inductor in series with the capacitor is connected in between the primary ground and the connection between the two primary windings of the two transformers. The AC signal across the primary windings is produced by a phase shift full bridge topology, depicted in FIG. 8A. This structure being another embodiment of the invention.  
         [0052]    [0052]FIG. 14A is a schematic diagram of an DC-DC Converter using the embodiment presented in FIG. 4A, The AC signal across the primary winding of the transformer is produced by two phase shifted two transistor forward topologies. This structure being another embodiment of the invention.  
         [0053]    [0053]FIG. 14B is a timing diagram of the circuit of FIG. 14A.  
         [0054]    [0054]FIG. 15 presents a magnetic-packaging structure suitable for the implementation of the embodiments of the invention. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0055]    For the AC-DC and DC-DC converters for high voltage application wherein the use of Schottky rectifiers is not possible due to high voltage across the rectifiers, one of the major obstacles is the reverse recovery loss of the rectifiers. The reverse recovery losses are proportional with the current slope through rectifier at turn OFF, the reverse voltage across the rectifier after the rectifier exhibits high impedance, the temperature, frequency of operation and the device characteristics. Additional snubber circuits are often used to reduce the voltage stress on the rectifiers during switching. The frequency of operation has to be also reduced which leads to poor volumetric efficiency of the converters. In FIG. 2 are presented a prior art an AC-DC converter. An AC source drives the primary of the transformer T 6 ,  950 . There is inherent leakage inductance in the transformer, which acts as a current source during commutation of the rectifiers  406  and  408 . In the right side of the rectifier there is another current source formed by  410 . During switching transitions there are two unmatched current sources on both ends of the rectifiers. This leads to high voltage spikes across the rectifiers  406  and  498 . Additional RC snubbers are necessary to maintain the voltage across the rectifiers below the breakdown voltage. This leads to higher power dissipation and high EMI.  
         [0056]    In FIG. 3A there is presented another prior art topology used for high voltage applications wherein an additional Lv.  12 , is added in the primary of the transformer Tr 1 . Additional clamping diodes D 5  and D 6  are also inserted. The reverse recovery losses in the secondary rectifiers are reduced due to controlled current slope during switching. The limitation of the current slope is due to Lv. The presence of two-unmatched current source such as the Lv and  30  leads also to voltage spikes across the secondary rectifiers  22 ,  28 ,  24  and  26 . This requires snubbers circuits across the output rectifiers, which will reduce the efficiency of the converter.  
         [0057]    This invention proposes several circuits wherein there is not need for snubber circuits across the secondary rectifiers, and the reverse recovery losses are minimized and under certain conditions eliminated. The maximum voltage across the rectifiers in this invention is clamped to the output voltage. This will allow a very efficient operation at high frequency in high power and high voltage applications leading to an increase of the power density of the converter.  
         [0058]    In FIG. 4A is depicted an AC voltage source Vs,  46 . The voltage signal produced by  46  is presented on time diagram  60  in FIG. 4B. From To to T 1  the voltage source  46  provides a positive voltage,  154 . The duration of this signal is referred as Ton. The Ton is the difference between T 1  and To. After T 1  the voltage source  46 , provides a zero amplitude voltage signal  156 , exhibiting in the same time very low internal impedance. During the time interval T 1 -T 2  , the voltage source  46  acts as a sort circuit. The current can flow bydirectional through  46 , without developing a significant voltage drop. After T 3 , the voltage source  46 , provides a voltage  158  of opposite polarity to  154  and the same amplitude. The time interval T 3  to T 4  when  46 , provides  158 , is equal to the time interval To to T 1 . The product of the time interval (T 1 -To) and the voltage amplitude  154  is equal to the product of the time (T 4 -T 3 ) and the voltage amplitude of  158 . After T 4 , the voltage source  46 , is equivalent again to a short circuit providing a zero amplitude voltage. This state will apply for the time interval between T 4  to T 6 . The time interval T 1  to T 3  and the time interval T 4  to T 6  are preferred to be equal, though it is not necessary. The embodiment of this invention will also apply if T 3 -T 1  is not identical to T 6 -T 4 .  
         [0059]    This type of voltage source can be produced by a transformer and several controlled switchers as is depicted in FIG. 8A, 9A and  14 A. The method of control pictured in  8 A is well known in the prior art.  
         [0060]    As is depicted in FIG. 4A in addition to the voltage source  46 , there is an inductive element Lr,  48  and a bridge of rectifiers,  900 , formed by rectifiers  50 ,  54 ,  52 , and  56 . The inductive element Lr is referred in this application as the soft commutation inductor element. The voltage source  46  is connected in series with inductor  48 , and applied across the rectifier bridge  900  with one termination to the cathode of  56  and the anode of  50 . The second termination goes to the cathode of  52  and the anode of  54 . A capacitor  58  is connected across the other two termination of the bridge, respectively between the anodes of  56  and  52  and the cathodes of  50  and  54 . A load  902  is connected across the capacitor  58 .  
         [0061]    During To to T 1  when  46 , produces a positive voltage  154  at the end connected to Lr, the current fill flow through Lr, rectifier  50 , rectifier  52  the capacitor  58  and the load  902 . It is assumed that the voltage across  58  exhibits low voltage ripple much lower than the voltage amplitude produced by  46 . With this assumption the current through Lr will build up linearly as pictured in  62  of FIG. 4B, from zero to a peak value at T 1 . During this period of time the energy provided by  46 , will be transferred to  902 , and a part of energy will remain stored in Lr. During the time interval between T 1  to T 2  when the voltage source Vs is equivalent to a short circuit, the current through Lr will continue to flow until the entire energy stored in Lr will be transferred to the load  902 . This will occur at T 2 , when the current through Lr will reach zero. During the interval T 2  to T 3  there is not energy transferred to the load. Do 1  and Do 1 ′ ceases to conduct after the current through Lr reaches zero. A voltage ringing may occur across the rectifiers of bridge  900 , as a result of the resonant circuit formed by Lr and the parasitic capacitance of the rectifiers,  50 ,  52 ,  54  and  56 . The energy contained in this resonant circuit is very small, if the parasitic capacitance of the rectifiers is small. A saturable inductor element is in series with Lr may damp this oscillation if its energy content is too high. In the critical conduction mode of operation wherein the voltage produced by Vs will change polarity after the current through Lr reaches zero this ringing will not exist. The time interval between T 2  and T 3  can be controlled in such way that the polarity change of the voltage produced by  46  will occur after the ringing across the rectifiers forces the second set the rectifiers, which will conduct in the next cycle, in this case  56  and  54 , in conduction. In this mode of operation there will be zero voltage switching for the rectifiers. The disadvantage of this mode of operation is the frequency modulation for load and line change. The advantage of this mode of operation is that we create ideal switching condition for the rectifiers. A combination of critical conduction through Lr and fix operating frequency for light load operation it may give the optimum operating mode. At the moment T 3 , the voltage source  46 , changes its polarity and the current fill start flowing linearly through Lr in the opposite direction. At the moment T 4 , when the voltage produced by  46 , becomes zero, the current through Lr reaches its peak. Further between T 4  to T 5  the entire energy stored in Lr will be transferred to the load  902 . This will occur at the moment T 5  when the current through Lr reaches zero. In between T 5  and T 6  we have the same mode of operation as described between T 2  and T 3 .  
         [0062]    The advantage of this mode of energy transfer from the source  46 , to the load  902  is the fact that the voltage across all the rectifiers contained in the bridge  900  is clamped to the voltage across Co which is also the output voltage of the converter and the fact that the current through the rectifiers at turn OFF is zero. The voltage across all the rectifiers reverses only after the current reaches zero. This mode of operations eliminates reverse recovery losses, which is a significant advantage for high output voltage application. If the mode of operation is critical conduction through Lr, and the voltage commutation of  46 , occurs after the current through Lr reaches zero and delayed by a time interval until the voltage across the rectifiers which will conduct at next cycle to reach zero, or its lowest level. The control of the voltage reversal of the voltage source  46  can be done through anagogic circuits which sense the current through Lr and the voltage across the rectifiers, or can be done with microcontroller using digital signal processing.  
         [0063]    In FIG. 6A there is presented the same configuration as FIG. 4A. The difference is the mode of operation, which is depicted in FIG. 6B. After a positive voltage is applied by  46 , and the current is buit up through Lr, until the voltage source  46 , becomes a short circuit. The current will continue to flow through Lr transferring its stored energy to the load. When the voltage produced by Vs changes its polarity there is current present through the Lr. The voltage across Lr is the sum between the output voltage  904  and the voltage across Vs,  158 . The current through Lr will decrease at a higher rate and will reach zero at T 8 . Between T 3  and T 8  the current will force the rectifier  50  and  52  to continue to conduct. The reverse of conduction will occur at T 8 , when the rectifier  56  and  54  will be forced into conduction. In this mode of operation there are reverse recovery losses due to the conduction of the rectifiers when the reverse voltage is applied to them. The reverse recovery losses are not high due to the fact that the rate of current change through rectifiers at turn off is limited by Lr. In addition to that the reverse voltage across the rectifiers is clamped to the output voltage  904 . During the time interval T 3  and T 8  and T 6  to T 9 , the energy is transferred from the Lr to the voltage source. In conclusion during T 7  to T 1  and respectively T 8  and T 4  the energy is transferred from the primary to the load and to the Lr. During the time interval To to T 7  and respectively T 3  and T 8 , the energy is transferred from Lr to the source. This kind of operation is less efficient that the one presented in FIG. 4A. In most of applications this mode of operation referred as continuous conduction mode can occur during the transient situations. The current through the Lr is depicted in  162 . The voltage across the rectifiers is depicted in  164  and  168 . The current through  50  and  52  is depicted in  166 . In  170  is depicted the current through  56  and  54 .  
         [0064]    It is very important to underline the fact that the key advantage of this invention is the energy transferred from primary to secondary is done in one direction only, if the operation occurs in discontinuous mode. The voltage source Vs would not exhibit short circuit characteristics during T 1  to T 3  and T 4  to T 5 , the energy will be also transferred from the load to the primary. This limits the numbers of topologies, which can be used in the primary side, capable to provide the right characteristics for Vs.  
         [0065]    In the circuit presented in FIG. 5A and 6A the energy transferred from the Vs to the load is function of the voltage across Vs, the output voltage  904  and the inductor Lr. The circuit  7 A presents a method of power transferred wherein the amount of energy transfer for a given Vs, Lr and Vo can be further increased and modulated by the additional bydirectional switch S 2 . The additional switch S 2  does not have to be bydirectional.  
         [0066]    The bydirectional switch will maintain the symmetry of the energy transfer for the positive cycle  154  and negative cycle  158 . A unidirectional switch will modulate the power only for positive or negative cycle. The timing diagram on FIG. 7B is referring to a bydirectional switch. The control signal for the switch S 2 ,  336 , is synchronized with the voltage source Vs. The switch is turned ON when a positive polarity voltage  154 , and a negative polarity voltage  158  is ramping up from Vs. During the time interval from To to T 10 , a positive voltage  154  is provided by Vs, and the switch  336  is ON. As a result the current will flow through Lr with a slope determined by the amplitude of  154  and the value of Lr. The value of the output voltage  904  does not play any role. The current slope in between To and T 10  is high and more energy is stored to Lr during this interval. When the switch S 2  is turned OFF at the moment T 10  the current which flows through Lr will turn ON the rectifiers  50  and  52 , transferring the energy to the load  902 . The current continues to build up through Lr at a smaller rate imposed by the difference between the amplitude of  154  and the output voltage. At the moment T 1  the current through Lr reaches its peak. From T 1  to T 2  all the energy contained in Lr will be transferred to the output if the unit operates in discontinuous mode, as is depicted in FIG. 7B. The unit can also operate in continuous mode, wherein the current will not reach zero before the voltage Vs will reverse its polarity. When the voltage Vs changes its polarity at T 3 , the switch S 2  will be also turned on and the current will build up through Lr with the same slope as between To to T 10 , but of opposite polarity. The major advantage of this embodiment over the embodiment presented in FIG. 4A and 6A, is the fact that the energy transfer from Vs to Lr is independent of Vo during the To to T 10 , respectively T 3  to T 12  period. The energy transferred during a given time is higher without increasing the amplitude of  154 . The output power can be also modulated not only by the ratio of the ON time which is the summation of the (T 1 -To) and (T 4 -T 3 ) and the period of the signal depicted in  160 . The output power can be further modulated by the ON time of S 2 .  
         [0067]    [0067]FIG. 10A is an extension of the embodiment depicted in FIG. 4A wherein the rectifiers  56  and  52  are replaced by the controlled synchronized rectifiers M 10 ,  802 , and M 11 ,  804 . These synchronous rectifiers are controlled by the Vc 2  and Vc 1  depicted in  700  and  702 . The use of synchronous rectifiers may have the advantage of lower voltage drop, which translates in a higher efficiency. Between To to T 1  when a positive voltage  154  is applied to Lr, the current will flow through Do 1  and M 11 . The control signal  908  turns ON the M 11  at To. The control signal  908  has to keep M 11 , turned ON until T 2 . The control signal Vc 1  can be maintain high even after T 2  without impacting the mode of operation. This is a major advantage for using synchronous rectifiers because the timing at turn OFF is not very critical. Important is to turn OFF, Vc 1  prior the T 3 .  
         [0068]    Another embodiment of this invention is presented in FIG. 11A. The bridge  900  is formed by the synchronous rectifiers controlled by controlled signals Vc 1  and Vc 2 . The timing diagram for Vc 1  and Vc 2  are depicted in FIG. 11B in  704  and  706 . The mode of operation for the converter depicted in FIG. 11A is the same as the one depicted in FIG. 4A for discontinuous and critical conduction mode and the one depicted on  6 A for the continuous conduction mode.  
         [0069]    Another embodiment of the invention is presented in FIG. 5A. The circuit formed by the voltage source Vs and the Lr in series is connected to a bridge formed by two rectifiers  134  and  136  and two capacitors  138  and  140 . The output voltage  904  is the voltage across the two capacitors  138  and  140 . The output voltage is further applied to a load. In this there are used only two rectifiers. Between To and T 1  the voltage source Vs provides a positive polarity voltage  154  at the end connected to Lr. The current will flow through Lr, Do 1  and Co 1 . At T 1  the current through Lr reaches its peak. Between T 1  to T 3  the source  46  becomes a short circuit  156 . The current continues to flow through Lr until reaches zero at T 2 . At that time all the energy stored in Lr is transferred to the Co 1  and Load, via Vo. At T 3  the voltage polarity produced by  46  reverses. The current will flow from the voltage source Vs through Co 2 , Do 2  and through Lr in an opposite direction to the flow in the previous cycle. The load is applied across the series combination formed by Co 1  and Co 2 . This circuit maintains the same advantages of the circuit depicted in FIG. 4A. The voltage across each rectifier is clamped to the output voltage. The current through rectifiers reaches zero prior to the application of a reverse voltage across them. As a result the reverse recovery losses are eliminated. If the converter operates in critical conduction mode, wherein the voltage polarity of  46  changes after the current through  48  reaches zero, with a delay necessary for the voltage across the rectifier which will conduct at next cycle reaches zero or close to zero, we can reach zero voltage or near zero voltage commutation for the rectifiers. The circuit depicted in FIG. 5A can also operate in continuous conduction mode as the circuit depicted in FIG. 6A. In FIG. 12A the rectifiers  134  and  136  are replaced by two controlled synchronous rectifiers  810  and  812 . The control signals Vc 1  and Vc 2  are depicted in FIG. 12B in  708  and  710 .  
         [0070]    In order to produce the Vs in the secondary of a transformer there are presented three circuits, which are suitable to produce such a source. One of these circuits is presented in FIG. 8A. The timing diagram associated with this circuit is presented in FIG. 8B. This circuit is familiar to those skilled in the art. It is known as phase shift full bridge. It is formed by two complementary half bridges, one formed by M 1  and M 2  and another one formed by M 3  and M 4 . The control signals for M 1  and M 4  are presented in  80 . The controlled signals for M 3  and M 4  are presented in  82 . During the conduction of M 1  and M 4  the input voltage is applied to the primary winding  110  of the transformer T 2 . The voltage induced in the secondary winding  112  is positive in reference to the arrow  116 . In the secondary winding  112  there will be a voltage referred previously as  154 . When the M 4  turns OFF the current will continue to flow through the primary winding  110 , and further through the body diode of M 3 , creating zero voltage switching condition for M 3  which is turned on at zero voltage. During the time when M 1  and M 3  conduct, the primary winding of  106  is shorted. In the secondary, the Vs will be zero and a short-circuit characteristics.  
         [0071]    This is equivalent to what previously was referred as  156 . At the moment when M 1  turns OFF the current will continue to flow through  110  and the drain to source capacitance of M 2  creating zero voltage or near zero voltage conditions for M 2 . The voltage applied to primary winding  110  will change the polarity applying a negative voltage in reference to the arrow  112 . This is equivalent to what previously was referred as  158 . When M 3  is turned OFF the current continue to flow through  110  discharging the drain to source capacitance of M 4  to zero or near zero, creating zero or near zero voltage switching conditions for M 4 . When M 2  and M 4  conduct the primary winding  110  is shorted and in the secondary the state of Vs is as short circuit  156 .  
         [0072]    The circuit presented in FIG. 8A , which is know as phase shifted full bridge converter can generate in the secondary of the transformer the voltage source used in describing our embodiments. What differentiate the circuit of FIG. 8A from other circuits which can generate a voltage source, is the short circuit behavior  156  during the time when the voltage in secondary is zero.  
         [0073]    The combinations of the circuits presented in FIG. 4A, 6A,  7 A  5 A,  10 A,  11 A and  12 A with the full bridge phase shifted topology depicted in  8 A, has another advantage. The slow rising the current through Lr in the secondary will allow the full swing towards zero voltage across all the switchers in the primary. A fast current ramp in the secondary winding which is specific to the prior art topologies as depicted in  2  and  3 A the soft switching in the primary is difficult to achieve for one of the complementary half bridge. This is due to the fact that the fast rise of the current in the secondary will steal some of the primary current flowing through the resonant tank formed by the magnetizing inductance of the transformer and the parasitic capacitance of the switchers.  
         [0074]    Another circuit capable to provide the secondary voltage Vs with the bydirectional low impedance characteristics during  156 , is depicted in FIG. 9A. This topology is not known by those skilled in the art. This topology is a modification of a conventional half bridge converter with the addition of a supplementary bydirectional auxiliary switch S 1 . The timing diagram is depicted in FIG. 9B. The switching elements  118  and  120  are controlled by the signals  122  and  124 . The control signals  122  and  124  have the same duration, in between these two signals is a dead time. By increasing the duration of  122  and  124  and accordingly decreasing the duration of the dead time, the power transferred to the output can be controlled. An additional control signal  132  controls the bydirectional switch S 1 . The control signal  132  is turning the switch  334  ON during the dead time  910 . There is a dead time between the falling edge of  122  and the rising edge of  132 . There is also a dead time between the falling edge of  132  and rising edge of  124 . This delay time is necessary to allow the voltage across the switching elements  118 ,  120  and  334  to swing in order to achieve zero voltage-switching conditions. In FIG. 9B as is depicted on  88 , the voltage across the switching element  120 , VM2, has a soft transition from Vin level to a voltage plateau Vin/ 2  during the conduction of SI and further a soft commutation to zero after the falling edge of  132 . The current through switching element  120  is depicted on  90 . During the conduction of  118  there is a voltage in the secondary winding  112  of the transformer T 2 . This is equivalent to  154 . During the conduction of  120  there is a negative voltage across  112  , equivalent to  158 . During the conduction of  132 , there is a short circuit across the primary winding  110 , which reflects in the secondary winding  112 . This state is equivalent to  156 .  
         [0075]    The topology described in FIG. 9A provides in the secondary winding  112  of the transformer  106  the voltage source with the characteristics required in our embodiments. In addition to this, the circuit of FIG. 9A offers zero voltage switching conditions for both switching elements, and recycles the leakage inductance energy which is not dissipate, but used for discharging the parasitic capacitance of  118 ,  120  and  334 .  
         [0076]    A third circuit capable to produce the required voltage source characteristics of Vs, is presented in FIG. 14A. There are two power trains, formed by two transistor forward topologies. The first power train contains two switching elements M 11  and M 12 , controlled by the same control signal Vc 11 ,  968 . The first power train contains also a transformer T 11 ,  988 . The second power train contains two switching elements M 13  and M 14 . Both switching elements are controlled by the same control signal, Vc 13 ,  972 . In the second power train there is a transformer T 12 ,  990 , which has the secondary winding  980  in series with the secondary winding  978  of the transformer  988 . The timing for Vc 11  and Vc 13  is presented in FIG. 14B, on  992  and  994 . The power is modulating by the phase shift between the Vc 11  and Vc 13 . The voltage in the secondary of the transformers  988  and  990  will substract during the overlapping time of Vc 11  and Vc 13 , creating in the secondary the  156  signal. In the secondary in series with the secondary windings  978  and  980 , there is the soft commutation inductive element Lr. The secondary rectifier means and the output filter is the one described in FIG. 4A.  
         [0077]    In FIG. 1A is depicted a circuit wherein the embodiment of claim  4 A is combined with the circuit described in FIG. 8A. There is an additional circuit formed by an inductor element  440  and a capacitor  442 , The additional circuit creates a triangular current waveform which is superimposed on the currents through M 1  and M 2 . In FIG. 1B is presented the timing diagrams of the key waveforms of the circuit illustrated in FIG. 1A. The control signals for M 1  and M 2  are presented on  914 . The control signal for M 3  and M 4  is presented on  916 . The triangular shaped additional current  922  flowing through  440  and  442  are presented on  918 . The current  924  flowing through M 1  is the result of the superposition of the  922  and the current reflected from the secondary of the transformer. The presence of  918  allows zero voltage switching conditions for M 1  and M 2 . The additional current  922  will add to the magnetizing current of transformer T 2  and discharge the parasitic capacitance of M 1  and M 2  prior the switchers M 1  and M 2  are turned ON. The magnitude of  922  is controlled by the size of  440 . A lower inductance of  440  will increase the additional current  922  . This will ensure the zero voltage switching conditions for M 1  and M 2 . The switchers M 3  and M 4  have an inherent zero voltage switching characteristics. If zero voltage switching has to be reached even at zero phase shift on both section of the full bridge, a similar circuit formed of an inductor in series with a capacitor can be inserted between the GND and the M 3  and M 4  at the node where the transformer T 2  is inserted. The voltage across the  442  and the additional capacitor is the same and equal to Vin/2. As a result the circuit can be simplified by connecting only one inductor with center tap across the primary winding  110 . The center tap of the additional inductor can be further connected to a capacitor which has the second termination connected to the GND. The capacitor  442  can be also formed by two capacitors in series one connected to the positive end of the Vin and the second capacitor connected to the negative end of  130 .  
         [0078]    The common node of these capacitors is connected to  440 . In FIG. 13A is presented the combination of the circuit presented in FIG. 4A and the full bridge phase shifted circuit depicted in FIG. 8A. The circuit presents another embodiment of the invention wherein the inductor element Lr  48  is transferred in the primary of the transformer T 2 . The mode of operation is similar with the circuit wherein the inductor element Lr is located in the secondary of the transformer. One advantage of this circuit is the fact that the current flowing through Lr will help to achieve zero voltage switching conditions for the primary switchers  92 ,  96 ,  94  and  160 , with the penalty of an increase in the flux density in the transformer&#39;s core  108 .  
         [0079]    In FIG. 13B the soft commutation inductor is split in two elements, one in the primary of the transformer  48 A and one in the secondary of the transformer  48 B. The ratio between  48 B and  48 A reflected in the secondary can be chosen for the optimization of the circuit. The optimization will be chosen for different criteria function of the priority of the design. It is important to understand that Lr, Lr 1  or Lr 2  can be implemented by the leakage inductance of the transformer. An additional discrete inductive element in series with the equivalent leakage inductance may or may not be necessary, function of the application.  
         [0080]    In FIG. 13C the soft commutation inductor element is split in two section  48 C and  48 D. These two sections are implemented on the same magnetic core  930 . An additional capacitor Czvs  932  is inserted in between the  48 C and  48 D and the ground. The same effect can be reached if the  932  will be connected to positive end of  130 .  
         [0081]    There are two transformers T 2  and T 600 , which have the primary windings  110  and  606  in series and the secondary windings  620  and  608  also in series. The invention does not limit to two transformers. It can be a number of transformers, preferable an even number and the connection to the capacitor  932  will be done in the middle having an equal number of transformers at each side of the connection.  
         [0082]    This circuit formed by  932 ,  48 C and  48 D will add supplementary currents, which will assist in achieving, zero voltage switching for  92 ,  96  ,  94  and  160 . When the diagonal switchers are conducting such as  94  and  96  or  92  and  160 , the combination LrC and LrD will exhibit a higher impedance calculated in a such way to achieve optimum energy transferred to the secondary as presented in FIG. 13A. When the upper switchers  92  and  94  or the lower switchers  96  and  104  conduct the impedance between the end of  932  not connected to the GND, and the transformers T 2  and T 600  primary winding is very small. This will lead to circulating currents through  932 , which will allow zero voltage switching conditions even at zero phase shift. This is very important in applications wherein zero voltage switching can be accomplished regardless of the phase shift.  
         [0083]    In FIG. 13D is depicted a circuit wherein the soft commutation inductor  48 , is transferred to the secondary, and the LrC and LrD is substituted by Lzv  440 . The combination  440  and  442  is connected in between the T 2  and T 600 . This circuit has the advantage of providing an additional triangular current through both sections of the full bridge, M 1  and M 2  and also M 3  and M 4 . This structure can offer zero voltage switching conditions on all four switching elements,  92 ,  96 ,  94  and  160  regardless of the phase shift, even at zero phase shift. This circuit offer significant advantages over the prior art, such as soft switching across the rectifiers  50 ,  52 ,  54  and  56 , and also across all the switching elements in the primary regardless of load, input voltage and phase shift.  
         [0084]    Many alterations and modifications may be made by those having ordinary skill in the art without departing from the spirit of the invention. For example, is the use of several transformers on each side of the connection between  440 ,  616 , and  606 . The capacitor  442  can be implemented by using two capacitors in series which have the non-common node connected to each end of Vin,  130 .  
         [0085]    In FIG. 15 is presented a packaging concept suitable with this invention. All the switching elements such as  200   a ,  200   b ,  200   c ,  200   d  and  214   a ,  214   b ,  214   c  and  214   d , are attached on a multilayers board  202 , and cooled by means of via or thermally conductive inserts located under the switching elements to a base plate  932  attached under the multilayers boards  202 . In between the base plate  932  and multilayers board  202  there is a thermally conductive insulation material  934 . The magnetic elements are constructed using spiral traces inside of the multilayer board  202  with cutouts  218 , to allow the magnetic cores  216  to penetrate through and to close the magnetic circuit with second magnetic core  936  attached from the bottom of the  202 . The thermally conductive plate is interrupted under the magnetic core or it can provide cavities to accommodate them. A supplementary soft elastic material  938  with good thermal conductivity is inserted in between the core and the metal plate. Some additional electronic components such as  210 ,  208  can also be placed on  202 . Pressed connectors such as  204   a ,  204   b ,  206   b  and  206   c  can be inserted in  202  to offer a low impedance path for the input and the output current to an external mother PCB. The advantage of this packaging is the reduction of the stray impedance associated with the interconnection between the switching elements and the magnetic elements. It offers also a solid mechanical construction suitable for demanding working environment conditions.  
         [0086]    The invention is defined by the following claims wherein may be substituted therein for obtaining substantially the same result even when not obtained by performing substantially the same function in substantially the same way.