Abstract:
A method for coupling power across a non-conducting membrane. A generator converts a DC source on a first side of a non-conducting membrane to a square-wave at a determined frequency. The generator output connects to a transformer and onto a first set of capacitor plates on the first side of the membrane. A second set of plates on the second side of the membrane form a set of coupling capacitors wherein the non-conducting dielectric membrane becomes part of the coupling-capacitor dielectric material. The second set of plates connects to a transformer and onto a non-linear circuit that converts the square-wave to DC voltage and current that can power a load such that the power delivered is approximately equal to the power available from the DC source on the first side of the membrane. The coupling capacitors may be replaced by coupled coils with nearly the same power delivery effect.

Description:
STATEMENT OF GOVERNMENT INTEREST 
   The invention described herein may be manufactured and used by or for the Government of the United States of America for governmental purposes without the payment of any royalties thereon or therefore. 

   CROSS REFERENCE TO OTHER PATENT APPLICATIONS 
   None. 
   BACKGROUND OF THE INVENTION 
   (1) Field of the Invention 
   The present invention relates to power transmission and more specifically to a method for coupling a direct current power source across a non-conducting membrane. 
   (2) Description of the Prior Art 
   Some “smart skin” systems use thousands of electronic sensors located on the external surface of an underwater vehicle in which the sensors maintain surveillance of the surrounding seawater. In some cases, it may be desirable to maintain the integrity of the vehicle skin as an impermeable membrane in that physical conduits to the sensors should not penetrate the membrane. As such, a need exists for a method of powering across a non-conducting membrane in order to provide primary power to the sensors. 
   SUMMARY OF THE INVENTION 
   Accordingly, it is a general purpose and primary object of the present invention to provide a method to provide primary power to a system on one side of a protective impermeable membrane that is derived from a power source located on an opposite side of the membrane. For example, to provide power to a system imbedded in a human body wherein the membrane is the human skin enclosing that body. 
   In order to attain the object described, the method efficiently couples a direct current (DC) power across a barrier (impermeable membrane) using circuits, which are realized with conventional electronic components. Placing parallel plates on opposite sides of the membrane can form coupling capacitors across the membrane. 
   For example, circular plates with a diameter of four centimeters and separated by a one-millimeter thick dielectric would have a capacitance of about ten pico-Farads multiplied by the relative dielectric constant of the material. A capacitance of this magnitude would support the disclosed method for many applications. As a result, the DC power is transmitted to the electronic circuits without physically penetrating the dielectric skin of the underwater vehicle, transmitting through a capacitive coupling or a magnetic coupling, which bridges the dielectric skin to power the sensors or other electric equipment. 
   To anyone or to those ordinarily skilled in the art will recognize that the method of the present application can also be realized by a dual method wherein a magnetic coupling mechanism is effected by placing coupled coils on topologically opposite sides of a non-magnetic membrane. While this application will focus on capacitive coupling in order to teach the method, the dual magnetically coupled mode is claimed implicitly. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     A more complete understanding of the invention and many of the attendant advantages thereto will be readily appreciated as the same becomes better understood by reference to the following detailed description when considered in conjunction with the accompanying drawings wherein like reference numerals and symbols designate identical or corresponding parts throughout the several views and wherein: 
       FIG. 1  depicts a Thevenin-Equivalent square-wave source; 
       FIG. 2  depicts a loaded square-wave source; 
       FIG. 3  depicts a square-wave source, positive half period; 
       FIG. 4  depicts a square-wave source, negative half period; 
       FIG. 5  depicts a square-wave generator; 
       FIG. 6  depicts a square-wave generator, steady state, OPEN switch; 
       FIG. 7  depicts a square-wave generator, steady state, CLOSED switch; 
       FIG. 8  depicts a square-wave generator, steady state energy exchange; 
       FIG. 9  depicts a [square-wave]-to-DC converter circuit; 
       FIG. 10  depicts a [square-wave]-to-DC converter, positive half period; 
       FIG. 11  depicts a [square-wave]-to-DC converter, negative half period; 
       FIG. 12  depicts a DC converter, steady state energy exchange; 
       FIG. 13  depicts a power transmission model; 
       FIG. 14  depicts a schematic for coupling a DC source across a dielectric membrane; and 
       FIG. 15  depicts a first variant of coupling a DC source across a membrane. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   A method for transmitting direct current (DC) power over a transmission line is disclosed in which the method uses two circuits. The method provides DC power to loads that have a barrier with respect to a source of the DC power. 
   The first circuit including a square-wave generator uses the DC power source to generate a square-wave. The generator uses an inductor, a capacitor, and a switch, which can be implemented using a single transistor and a drive circuit. The second circuit, a square-wave converter, converts a square-wave source into a DC source. The converter uses an inductor, a capacitor and a P-N junction diode. 
   The generator and converter each provide a matched termination to a uniform transmission line so that energy is not reflected from the converter back toward the generator. Thus, the transmission line can be lengthened without affecting the efficiency of power transmission. 
   The characteristic impedance of a “uniform” transmission line is constant over an entire length of the transmission line. Power will be lost and a voltage drop will occur as a result of transmission line Ohmic losses. Using transmission line transformers to increase voltage and decrease current on the transmission line can reduce these losses. 
   The transmission efficiency of the disclosed method is determined by departures from ideal parameters of the components used. If the diode, inductor, capacitor, and transmission line were all ideal and loss did not exist, then the transmission efficiency would be one hundred percent. 
   A small amount of DC power is required to energize the switch driver circuits but the switch driver is located at the DC source where, presumably, DC power is more than sufficient. Therefore, a discussion of the switch driver power is minimized relative to the method of the disclosure. That is, the DC power delivered to the load would be equal to the DC power available from the source. 
   There is a limit on the amount of power that can be delivered to a load using the disclosed method. The limit is fixed by the non-linear properties of the diode used in the converter circuit. The maximum current that can be delivered to a load is one half of the maximum forward current that the diode can safely carry. The maximum voltage that can be delivered to a load is one half of the breakdown voltage of the diode. The maximum DC power that can be delivered to a load is the product of one-half of the maximum forward current and one half of the breakdown voltage. For example, a diode with a reverse breakdown voltage of 100 Volts may be able to support a maximum forward current of one Ampere. The maximum power that could be delivered to a resistive load by a converter that uses this diode would be 25 Watts (50 Volts×500 mA). 
   The disclosed method uses inductors and capacitors as energy storage elements and it is well known that energy is lost when an abrupt change in capacitor voltage or an abrupt change in inductor current is required by a circuit operation. Within the disclosed method and under steady-state operating conditions, inductor current and capacitor voltage remain essentially constant. “Steady-state” operating conditions are the operating conditions under which the circuits would normally be used. When the circuits of the disclosed method are first energized and the switch begins operation; the current through the inductors and the voltage across the capacitors are both zero. A transient state exists until the inductor currents and capacitor voltages become periodically stable. These operating conditions are thus described as “steady-state”. 
   Under most conditions, inductor voltage and capacitor current are each subjected to abrupt changes as the square-wave polarity changes. Even though these abrupt changes are allowable with ideal components, the parasitic capacitance of the inductors and the parasitic inductance of the capacitors degrade the ideal operation of the method and decrease the observed efficiency. In this discussion, these parasitic effects are minimized because, in good-engineering practice, these effects only minimally degrade performance. 
   In  FIG. 1 , a Thevenin-Equivalent square-wave source  2  is illustrated. In the figure, a signal generator  4  switches a voltage between a positive value state, +V S  and a negative value state, −V S , which have the same magnitude but opposite polarity. In  FIG. 1  and the follow-on figures, E(t) refers to a voltage, E, that is a function of time, t. The time graph below the circuit describes the voltage-time function as a square wave. Thus, E(t) refers to the time-varying voltage illustrated in the bottom half of each figure. 
   The switching operation is periodic with a period, T S , and with equal dwell times in each state. Thus, the average value of the generator voltage is zero. Furthermore, the time, T S , that is required to switch between the two states, is negligible. 
   E(t) in each figure refers to a voltage, E, that is a function of time, t. The time graph below the circuit describes the voltage-time function as a square wave. Thus, E(t) refers to the time-varying voltage illustrated in the bottom half of each figure. 
   The source impedance  6  of the square-wave generator  4  is Z 0 , a positive real number. In general, a Thevenin-Equivalent source impedance can be complex and may, under certain circumstances, have a negative real part. However, for the purposes of this disclosure, only positive real values of Z 0  are considered. This restriction is consistent with almost all practical applications. 
   If the equivalent circuit were used to drive a transmission line with characteristic impedance also equal to Z 0 , then the equivalent circuit for the output of the transmission line would be identical to the illustration in  FIG. 1 , regardless of the length of the transmission line. 
   The maximum power available from the Thevenin-Equivalent circuit of  FIG. 1  is equal to the power that would be delivered to a load resistor equal to Z 0 , as illustrated in  FIG. 2 . In  FIG. 2 , maximum power transfer occurs when a generator is driving a load that is equal to the source impedance of the generator. A load  12  is Z 0 , which is equal to a generator impedance  14 . Under these conditions, the voltage across the load is one-half voltage of the generator and the current is one half of the short circuit current available from the Thevenin-Equivalent generator. 
   During the positive state of a generator  32 , illustrated in  FIG. 3 , a current  34  passing through a load, Z 0 , is V S /2Z 0  so that the instantaneous power delivered to the load, Z 0 , is (V S ) 2 /4Z 0 . During a positive half period  36  of the square-wave cycle, the current  34  is positive and equal to the peak voltage, V S , divided by the total circuit resistance 2Z 0  and a voltage  38  across the load is one-half peak voltage, V S , of the generator  32 . 
   During a negative state of the generator  32 , illustrated in  FIG. 4 , the instantaneous power delivered to the load, Z 0 , is the same, (V S ) 2 /4Z 0 , even though the current  34  flows in the opposite direction. Thus, the average power is equal to the instantaneous power and is defined as P MAX =(V S ) 2 /4Z 0 , which is the maximum power available from the source. During a negative half period  40  of the square-wave cycle, the current  34  is positive and equal to the peak voltage, V S , divided by the total circuit resistance, 2Z 0  and the voltage  38  across the load is one half of the peak voltage, V S . Thus terminated, the generator is optimally loaded because the generator is delivering a maximum available power to the load, Z 0 . 
   Energy Efficient Square-Wave Generator 
   The circuit illustrated in  FIG. 5  can convert a DC source  50  into a square-wave  52  driving a load impedance  54  which is equal to an internal impedance  56  of the DC source. Assuming ideal components, an inductor  57  and a capacitor  58 , the efficiency of the conversion is approximately one hundred percent because the average square-wave power delivered to the load impedance  54  is equal to the maximum DC power available from the DC source  50 . A square-wave is created by the periodic operation of a switch  59  that changes state once each period, T S . The two states of the switch  59  are defined as follows: (1) when the switch is OPEN, the current through the branch containing the switch is zero while the voltage across the branch can assume any value, and (2) when the switch is CLOSED, the voltage across the branch containing the switch is zero while the current through the branch can assume any value. The dwell time in each of the two switch states is the same. 
   The switch  59  opens and closes periodically causing a square-wave of current to pass through the load resistor, Z 0 . A transient state occurs when the action of the switch  59  is first initialized. The transient state lasts until the voltage across the capacitor  58  and the current through the inductor  57  each reach a steady-state condition. 
   An OPEN condition of the steady-state switch  59  is illustrated in  FIG. 6 . During the depicted half period, the switch  59  is OPEN so that the current through the switch branch is zero. A steady-state DC current  62  equal to V DC /2Z 0  passes through the load, Z 0 , producing a voltage, V DC /2. During this half period T S /2, energy is supplied to the circuit by the decay of flux linkages in the inductor  57  while additional energy is being stored by increasing the charge held by the capacitor  58 . The voltage across the OPEN switch  59  is V DC . The average, steady-state, energy stored on the inductor  57  is E AVG ={L(V DC ) 2 }/8 (Z 0 ) 2  in which “L” is the inductance of the inductor. The energy delivered to the circuit, E DEL , by the inductor  57  during each OPEN-SWITCH condition should be a small fraction of the average energy, E AVG . This will be true if the inductance is much greater than the product, Z 0 T S . Thus, it is required that L&gt;&gt;Z 0 T S . During the OPEN condition of the switch  59 , the voltage across the branch of the switch is V DC . 
   A steady state CLOSED condition of the switch  59  is illustrated in  FIG. 7 . During the depicted half period, the switch  59  is CLOSED so that the voltage across the switch branch is zero. A steady state DC current, I DC , equal to V DC /2Z 0  reverses through the load, Z 0 , producing a voltage, −V DC /2. During this half period, T S /2, energy is supplied to the circuit by the capacitor  58  while new energy from the source is being stored in the inductor  57 . The current through the CLOSED switch  59  is V DC /Z 0 , which is twice the steady state DC current supplied by the DC source  50 . The average, steady state, energy stored on the capacitor  58  is E AVG =C (V DC ) 2 /8 in which “C” is the capacitance of the capacitor. The energy delivered to the circuit by the capacitor  58  during each CLOSED condition half period is E DEL =T S (V DC )  2 /8Z 0 , which is the same as that delivered by the inductor  57  during each OPEN condition half period. The choice of value of the capacitor  58  is made by observing that the delivered energy, E DEL , should be a small fraction of the average energy, E AVG . This will be true if the capacitance is much greater than the ratio T S /Z 0 . Thus, C&gt;&gt;T S /Z 0  is required. During the CLOSED condition of the switch  59 , the current through the branch of the switch is V DC /Z 0 , which is twice the current from the DC source  50 . 
   Energy balance is achieved if the ratio of the element values, L and C, are chosen such that (L/C)=(Z 0 ) 2 . The average energy stored on each element is the same. The energy exchange during each period of steady state operation is illustrated in  FIG. 8 . 
     FIG. 8  depicts the time variation of the energy stored on the inductor  57  and capacitor  58  in the square-wave generator illustrated in  FIG. 5 ,  FIG. 6  and  FIG. 7 . During each half period, energy is delivered to the circuit by either the inductor  57  or the capacitor  58  while the energy stored on the other component is increasing. During the next half period, the process reverses. The figure is based on an assumed condition that E DEL &lt;&lt;E AVG . Only one period is illustrated because, in the steady state, each period is identical to every other period. 
   By comparing  FIG. 6  with  FIG. 7 , the effects caused by the switch CLOSING operation are shown. When the switch  59  closes, the voltage across the inductor  57  changes polarity, but not magnitude, while the current through the capacitor  58  and a load impedance  70  changes direction, but not magnitude. Both of these instantaneous changes are permissible by the boundary conditions imposed by the circuit components and no transient behavior occurs as a result of the CLOSING operation of the switch  59 . The current through the inductor  57  and the voltage across the capacitor  58  do not change when the switch  59  CLOSES and this is also required by the respective boundary conditions of the inductor and the capacitor. 
   In one embodiment, the switch  59  used to implement the square-wave generator circuit could be a transistor collector emitter circuit. A small amount of energy is necessary to power a switch driver to provide the base-emitter drive current, which can be more than hundred times less than the peak collector-emitter current, V DC /Z 0 , when the switch is CLOSED. 
   Energy Efficient [Square-Wave]-to-DC Converter 
   In  FIG. 9 , an energy efficient [square-wave]-to-DC converter circuit is illustrated. The passive circuit requires a capacitor and an inductor for energy exchange and a single diode. A square-wave source  90  switches periodically between a positive voltage, +V S  and a negative voltage, −V S . After a steady state condition is reached, an inductor  92  acts as a constant current source delivering a positive DC current to the load resistor, Z 0 . 
   A transient state occurs when the square-wave source  90  is first initialized. The transient state lasts until the voltage across a capacitor  93  and the current through the inductor  92  each reach a steady-state condition. The square-wave source  90  and a source impedance  94  represent the Thevenin-Equivalent of a transmission line being driven by a square-wave generator. If the inductor  92 , the capacitor  93 , and a diode  95  of the converter circuit are assumed to be ideal, then the efficiency of the converter circuit is one hundred percent. That is, the DC power delivered to a load resistor  96  is equal to the maximum power available from a Thevenin-equivalent generator. 
   The square-wave is converted to DC by a non-linear property of the diode  95  that, in one state, permits an undefined current to flow through a branch of the diode in only one direction while the voltage across the branch containing the diode is zero and that, in a second state, permits an undefined voltage across the branch of the diode in only one polarity while the current through the branch of the diode is zero. 
   The operation of the converter circuit in steady state is described by observing each non-linear state separately. When the diode polarity is as illustrated in  FIG. 9 , the two states correspond to the NEGATIVE half period of the square-wave and to the POSITIVE half period of the square-wave, respectively. 
   The converter circuit steady-state operation during the POSITIVE half period is illustrated in  FIG. 10 . During the depicted half period, the square-wave source  90  (generator) presents a POSITIVE voltage, V s , to the circuit causing a current, V s /2Z 0 , to flow in the circuit. The diode  95  is reverse-biased by a voltage equal to V s  so that no current flows in the branch containing the diode. Thus, the current I s  flows through the load Z 0 , thereby generating a voltage V s /2 across the load. During this half period, the capacitor  93  supplies energy to the circuit while the inductor  92  is storing energy. The current through a branch of the diode  95  is zero. 
   The current driven by the square-wave source  90  flows through the capacitor  93 , the inductor  92 , and the load resistor  96 . During this half period, energy is delivered to the circuit by the capacitor  93  while the stored energy of the inductor  92  is increasing. The average, steady state, energy stored on the capacitor  93  is E AVG =C (V S )  2 /8 in which “C” is the capacitance of the capacitor. The energy delivered to the circuit by the capacitor  93  during each positive half period is E DEL =T S  (V S ) 2 /8Z 0 . The choice of value of the capacitor  93  is made by observing that the delivered energy, E DEL , should be a fraction of the average energy, E AVG . This will be true if the capacitance is much greater than the ratio, T S /Z 0 . Thus, it is required that C&gt;&gt;T S /Z 0 . During the POSITIVE half periods of the square-wave, the voltage across the branch of the diode  95  is V S  with a polarity that reverse-biases the diode junction so that no current can flow in the branch containing the diode. 
   The converter circuit steady state operation during a NEGATIVE half period is illustrated in  FIG. 11 . During this half period, the square wave source  90  presents a negative voltage-V s  to the circuit causing a current −V s /2Z 0  to flow in the circuit. The diode  95  is forward-biased by a current equal to V s /Z 0  and the voltage across the branch containing the diode is about zero. A current I S  flows through the load Z 0 , generating a voltage V s /2 across the load. During the depicted half period, the capacitor  93  is storing energy while the inductor  92  supplies energy to the circuit. 
   The voltage across the branch of the diode  95  is zero and the current through a branch of the diode is V S /Z 0 , which is twice a current  100  driven by the square-wave source  90 . During this half period, T S /2, energy is delivered to the circuit by the inductor  92  while stored energy of the capacitor  92  is increasing. An average, steady state, energy stored on the inductor  92  is E AVG ={L(V S ) 2 }/8(Z 0 ) 2  in which “L” is the inductance of the inductor. The energy delivered to the circuit by the inductor  92  during each NEGATIVE half period is E DEL =T S  (V S ) 2 /8Z 0 . The choice of value of the inductor  92  is made by observing that the delivered energy, E DEL , should be a small fraction of the average energy, E AVG . This will be true if the inductance is much greater than the product Z 0 T S . Thus, it is required that L&gt;&gt;Z 0 T S . 
   Energy balance is achieved if the ratio of the element values, L and C, are chosen such that (L/C)=(Z 0 ) 2 . Then, the average energy stored on each element is the same. The energy exchange during each period of steady state operation is illustrated in  FIG. 12 . 
     FIG. 12  depicts the time variation of the energy stored on the inductor and capacitor in the DC converter illustrated in  FIG. 9 ,  FIG. 10  and  FIG. 11 . During each half period, energy is delivered to the circuit by either the inductor or the capacitor while the energy stored on the other component is increasing. During the next half period, the process reverses.  FIG. 12  is based on an assumed condition that E DEL &lt;&lt;E AVG . Only one period is illustrated because, in the steady state, each period is identical to every other period. 
   By comparing  FIG. 10  with  FIG. 11 , the effects caused by the instantaneous change in square-wave polarity from POSITIVE to NEGATIVE are shown. When a square-wave polarity  101  changes, the voltage across the inductor  92  also changes polarity, but not magnitude, while a current  102  through the capacitor  94  changes direction, but not magnitude. Both of these changes are permissible by the boundary conditions imposed by the circuit components and no transient behavior occurs as a result of the polarity change. The current through the inductor  92  and the voltage across the capacitor  93  does not change when the square-wave polarity changes. The current through the square-wave source  90  and a load  103  is the same as the current through the inductor  92  and does not change in either polarity or magnitude. Thus and as predicted, the load resistor  96  experiences direct current (DC). 
   By comparing  FIG. 10  with  FIG. 3  and  FIG. 11  with  FIG. 4 , the converter circuit is indistinguishable from a resistive termination, Z 0 , when the source voltage is a square wave. 
   Consider a boundary  105  shown in  FIG. 10  and a boundary  39  shown in  FIG. 3 . The square-wave source  90  (generator) to the left of the boundary  105  in  FIG. 10  is identical to the generator  32  to the left of the boundary  39  in  FIG. 3 . 
   The current  100  crossing the boundary  105 , from the square-wave source  90  to the converter circuit is identical to the current  34  crossing the boundary  39  from the generator to the matched termination, Z 0 . The voltage across the boundary  105  is V S /2, which is identical to the voltage across the boundary  39 . Thus, it can be concluded that, during the POSITIVE half cycle of a square-wave  35  and a square-wave  106 , the converter circuit is indistinguishable from a resistor having a value, Z 0 . 
   Consider the boundary  105  shown in  FIG. 11  and the boundary  39  shown in  FIG. 4 . The square-wave source  90  and a Thevenin-Equivalent circuit  110  to the left of the boundary  105  in  FIG. 11  is identical to the current  34  and a Thevenin-equivalent circuit to the left of the boundary  39  in  FIG. 4 . 
   The current  102  of the capacitor  93  crossing the boundary  105 , from the converter circuit to the Thevenin-Equivalent generator is identical to the current  34  crossing the boundary  39  from the matched termination, Z 0 . The voltage across the boundary  105  is V S /2, which is identical to the voltage across the boundary  39 . 
   Thus, it can be concluded that, during the negative half cycle of the square-wave  35  and  101 , the converter circuit is indistinguishable from a resistor having a value, Z 0 . 
   After reaching a steady state condition, the disclosed converter circuit is indistinguishable from a resistive termination Z 0 , when driven by a square-wave. This property allows the converter circuit to be used as a matched termination for a uniform transmission line of any length with a transmission line characteristic impedance equal to Z 0  when the uniform transmission line is driven by a square-wave source. This property is illustrated in  FIG. 13 . 
   In  FIG. 13 , power from a DC source  130  is transmitted over a distance L T  (transmission line length), using an energy efficient square-wave generator  131  and an energy efficient [square-wave]-to-DC converter  132 . The source impedance  133  of the DC source, the characteristic impedance of transmission line  134  and the DC load impedance are each equal to Z 0 . 
   The transmission line  134  can be comparatively long since the [square-wave]-to-DC converter  132  presents a matched termination to the transmission line. The matched termination ensures that there are no reflections or standing waves on the transmission line  134  that would corrupt the operation of the system. 
   Method for Coupling DC Power Across An Impermeable Non-Conducting Membrane 
   In  FIG. 14 , power transmission through coupling capacitors  140  can provide power to electrical circuits inside a sealed environment. In the figure, broadband transmission line transformers  141  and  142  are used to step up the impedance level by a factor of “K” in the case of the capacitive coupling or step down by a factor of “p” in the case of an inductive coupling. Thus, the time constant of the coupling-capacitor circuit is increased by the factor K. A well-known property of transmission line transformers is that the factor K or the factor “p” can be any ratio of the form (M 2 /N 2 ) in which “M” and “N” are integers. In practice, impedance levels in broadband circuits can range between a low of about 6 Ohms and a high of about 600 Ohms. The largest practical value for KZ 0  is typically less than 600 Ohms. 
   A transmission line  143  that is coupled across a dielectric membrane using the two coupling capacitors  140  is illustrated. The wideband transmission line transformers  141  and  142  raise the impedance level at the dielectric boundary by the factor K in order to increase the transmission efficiency across the dielectric boundary. Wideband transmission-line transformers are extensively used in circuit applications covering a few MHz to a few GHz. Designs have been described and are known to those skilled in the art covering transformation ratios to approximately 64:1. In general, any impedance ratio of the form (M 2 /N 2 ), in which M and N are integers, can be realized using the wideband transmission line concepts. 
   In principal, the impedance level KZ 0  can be comparatively very large. However, practical circuit limitations will usually constrain KZ 0  to a maximum of about 600 Ohms. Assuming that the coupling capacitor  140  is 10 pico-Farad (pF), the period T S  of a square-wave generator  144  will be constrained by T S &lt;2RC, or T S &lt;2[(600)×(10 −11 )], or T S &lt;12 nano-seconds. Thus, the frequency of the square-wave would be about 50 MHz, which is within the practical limitations of the disclosed method. 
   The square-wave generator  144  converts a DC source  145  with an internal impedance, Z 0  (load resistor  146 ) to a zero average value square-wave source at an impedance level, Z 0  (load resistor  147 ). The output of the square-wave generator  144  drives the wideband transmission-line transformer that raises the impedance level of the square-wave source to KZ 0 . The output of the transmission line transformer  141  is capacitor-coupled across a dielectric membrane  200  (or other impermeable membrane) to the second transmission line transformer  142  that reduces the impedance level of the square-wave to Z 0 . The output of the second transmission-line transformer  141  feeds a converter  149  that converts the square-wave to a direct current, which feeds the load resistor  147  with an impedance, Z 0 . 
   In a variant of the method of the present application, a dual coupling mode may be accomplished. The method will be recognizable to those ordinarily skilled in the art by recognizing that the method of the present application can also be realized by a dual method wherein a magnetic coupling mechanism is effected by placing coupled coils  300  and  310  on topologically opposite sides of a non-magnetic membrane  400 . 
   While this application focused on capacitive coupling in order to teach the method, the dual magnetically coupled mode is claimed implicitly and supported by  FIG. 15 . In the former case, the coupling takes place at a high impedance level by means of coupling capacitors, whereas, in the dual case, the coupling takes place at a low impedance level by the coupled coils  300  and  310  on opposite sides of the non-magnetic membrane  400 . In the capacitor coupling case, the transformer impedance ration is labeled “K” whereas in the dual inductive coupling case, the transformer ratio is labeled “p” in order to differentiate the two impedance ratios. 
   In the dual case of the non-magnetic membrane  400 , the transformer  141  is designed to convert the source impedance to a low value (6 Ohms for example) because the magnetic coupling depends on current (Ampere turns). One coil  300  is placed on the DC source side while a second coupled coil  310  is placed on the load side. The non-magnetic membrane  400  topologically separates the two coils  300 ,  310  so that the spatial integrity of the load container can be maintained. The two-terminal outputs of the transformer  141  are electrically connected to two terminals  302 ,  304  of the first coupled coil  300  and two-terminal outputs  312 ,  314  of the second coupled coil  310  are electrically connected to the terminal output of the second transformer  142 . 
   A non-magnetic membrane may be necessary in some applications in order to provide an electric-field shield for an electronic system. One example of a non-magnetic membrane is aluminum foil. 
   It will be understood that many additional changes in the details, materials, steps and arrangement of parts, which have been herein described and illustrated in order to explain the nature of the invention, may be made by those skilled in the art within the principle and scope of the invention as expressed in the appended claims.