Abstract:
Bidirectional power conversion systems provide the ability to change power attributes to and from a component. Current bidirectional power conversion systems use a unidirectional power converter for each direction. The integration of the two normally independent power converters results in a bidirectional power converter with nearly half the size, weight, volume, cost and complexity. Described are embodiments of bidirectional power conversion systems that allow power transfer between two or more components without requiring the use of separate unidirectional power converters.

Description:
RELATED APPLICATIONS 
   This application is a Divisional of U.S. application Ser. No. 11/114,267, filed Apr. 25, 2005, now U.S. Pat. No. 7,348,767 which is a Divisional of U.S. Application No. 10/270,799, filed Oct. 11, 2002 now U.S. Pat. No. 6,894,461, the entire contents of each of which are hereby incorporated by reference. 

   BACKGROUND OF THE INVENTION 
   This invention relates to systems and methods for producing bidirectional power conversion. More specifically, this invention relates to circuits and methods for creating a truly bidirectional power converter with multiple control loops and power routing. 
   As the need for smaller and more intricate integrated circuits surfaces, so does the need for smaller power routing and power conversion processes. Power routing and power conversion processes may be found, for example, in uninterruptible power systems, next-generation automotive electrical systems, redundant battery systems for computers, telecommunications equipment and portable electronic devices. 
   Bidirectional power conversion provides the ability to change power attributes to and from a component. Bidirectional power conversion may be found in portable devices such as cell phones and Personal Digital Assistants (PDAs) that contain rechargeable batteries. These DC rechargeable batteries are automatically charged when connected to an AC power source. During battery charging, the power from the AC power source is converted into DC power (either externally with a wall adapter or within the portable device circuitry), and the battery is charged. Additionally, during battery charging, the portable device operates using the power supplied from the AC power source. When the portable device is not connected to the AC power source or wall adapter, the battery serves as the primary power source. This process of supplying power to and receiving power from the battery is in accordance with the principles of bidirectional power conversion. Furthermore, the process of routing power from the DC battery instead of the AC power source or wall adapter is present in such systems. 
   In prior systems, truly bidirectional power conversion is not realized. Instead, prior systems perform bidirectional power conversion through two separate unidirectional power converters and an additional power routing system to switch between the two converters. Each separate unidirectional power converter is dedicated to the power conversion process associated with a particular direction. The result of using multiple power converters in addition to additional logic for the purpose of routing in prior systems is that the amount of switches present is inopportunely large. 
   In light of the foregoing, it would be desirable to provide a truly bidirectional power converter with multiple control loops and power routing in order to reduce the complexity, size and cost associated with current converters that utilize multiple unidirectional power converters. 
   SUMMARY OF THE INVENTION 
   The present invention provides systems and methods for providing bidirectional power conversion using a single converter and controller for power routing. Through the use of only a single converter and controller, the amount of switches used in fabricating the circuit of the present invention is significantly reduced. In fact, the number of necessary switches may be nearly halved. Moreover, the volume, cost, complexity, and size of a bidirectional power converter in accordance with the principles of the present invention may also be nearly halved. 
   In addition to the above, further consideration relating to bidirectional power conversion should preferably also be given to the need for relatively high efficiency, which is measured as a ratio of the power output to the power input. Higher efficiency can be achieved by reducing system losses, and can enable extended battery operating time. The present invention therefore combines power routing and power conversion, and in doing so, eliminates the necessity for several switches and reduces total power (I 2 R) losses. Furthermore, the power supply switching techniques used in accordance with the principles of the present invention provide additional efficiency in bidirectional power conversion systems. 
   Moreover, although the following methods and systems to be presented assume external AC-DC conversion (e.g., power from an AC power source is converted externally by a wall adapter to supply DC power to be used in the bidirectional power converter), the present invention is not limited in this manner. AC-DC and DC-AC converters are readily available and may be incorporated into the embodiments of the present invention, though they have been excluded from the embodiments described in applicants&#39; specification in order to avoid the inclusion of material that would unnecessarily complicate the description of the present invention. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The above-mentioned objects and features of the present invention will be more clearly understood from the following detailed description considered in conjunction with the following drawings, in which the same reference numerals denote the same structural elements throughout, and in which: 
       FIG. 1  is a block diagram of a bidirectional power converter in accordance with the principles of the present invention. 
       FIG. 2  is a circuit diagram of a bidirectional buck and boost converter in accordance with the principles of the present invention. 
       FIG. 3  is a circuit diagram of a bidirectional buck and boost converter with additional power buses in accordance with the principles of the present invention. 
       FIG. 4  is a circuit diagram of a bidirectional flyback converter in accordance with the principles of the present invention. 
       FIG. 5  is a circuit diagram of a bidirectional flyback converter with additional power buses in accordance with the principles of the present invention. 
       FIG. 6  is a circuit diagram of an error amplifier with sense signal and compensation selection in accordance with the principles of the present invention. 
       FIG. 7  is a circuit diagram of a portion of a controller for a current-mode bidirectional power circuit in accordance with the principles of the present invention. 
       FIG. 8  is a circuit diagram of a preferred embodiment of a simplified current-mode buck battery charger and boost regulator power circuit in accordance with the principles of the present invention. 
       FIG. 9  is a circuit diagram of another embodiment of a simplified current-mode buck battery charger and boost regulator power circuit with the principles of the present invention. 
       FIG. 10  is a circuit diagram of a preferred embodiment of a simplified bidirectional flyback converter in accordance with the principles of the present invention. 
       FIG. 11  is a circuit diagram of another embodiment of a simplified flyback converter in accordance with the principles of the present invention. 
       FIG. 12  is a circuit diagram of a driver circuit used to control power transfer in a bidirectional power converter in accordance with the principles of the present invention. 
       FIG. 13  is a circuit diagram of a simplified adaptive slope compensation network in accordance with the principles of the present invention. 
       FIG. 14  is a circuit diagram of an embodiment of an adaptive slope compensation network for a bidirectional power converter in accordance with the principles of the present invention. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIG. 1  shows a simplified schematic  100  of a bidirectional power converter in accordance with the principles of the present invention. Schematic  100  comprises controller  110 , reactive elements  120 , switches S 0   131 , S 1   132 , SN  133  and SGND  134  and power buses B 0   141 , B 1   142  and BN  143 . In schematic  100 , as well as in the following circuits to be presented in accordance with the principles of the present invention, the switches shown are generally single-pole, single-throw switches. In practice, these switches may be realized, for example, by using semiconductor devices such as power metal-oxide semiconductor field effect transistors (MOSFETS), insulated gate bipolar transistors (IGBTs), bipolar junction transistors (BJTS), thyristors or other suitable devices. 
   Switch S 0   131  of  FIG. 1  is used to control the flow of power between bus B 0   141  and reactive elements  120 . Similarly, switches S 1   132  and SN  133  control the power flow between bus B 1   142  and reactive elements  120 , and between bus BN  143  and reactive elements  120 , respectively. Moreover, as shown in  FIG. 1 , switch SGND  134  controls power flow between reactive elements  120  and ground. 
   Reactive elements  120  may be one or more components in which power may be supplied to (and stored) as well as taken from. For example, reactive elements  120  may be any combination of inductors, transformers or capacitors. The invention is not limited in this manner. 
   In accordance with the principles of the present invention, switches S 0   131  to SN  133  may perform up to four functions. As explained in more detail below, these functions include charging reactive elements  120 , providing synchronous rectification, discharging reactive elements  120  (to drop any stored charge), and providing power routing. 
   Moreover, although switches are shown to couple each of the power buses to the reactive elements in  FIG. 1  and the figures to follow, other types of circuit components which are suitable for a particular application may be used. For example, assuming that power bus B 1   142  in  FIG. 1  is a unidirectional power port (i.e., it is only intended to either receive power from or transfer power to the reactive elements, but not both), a diode may be used in place of switch S 1   132 . Accordingly, the remaining power buses (using switches to coupled to the reactive elements) are able to transfer power bidirectionally with the reactive elements, but power bus B 1   142 , using only a single diode for coupling to the reactive elements, can only transfer power in one direction. The use of diodes for this purpose, however, can present problems for low voltage applications due to the voltage drops associated with the diodes. Accordingly, although diodes and other types of suitable components may be used, either alone or in combination (e.g., the use of multiple components to allow power flow in dual directions), the use of switches presents the simplest implementation of the present invention. 
   Controller  110  in  FIG. 1  uses current measurements of reactive elements  120  to monitor the direction and magnitude of current flowing through reactive elements  120  (shown as Isense bus  111 ). Although a single bus is shown in  FIG. 1 , however, it should be understood that more than one current sense signal may be used by controller  110  (i.e., a separate sense signal associated with each or any number of power buses can be utilized by controller  110 ). Using Isense  111 , controller  110  opens and closes switches S 0 -SN in order to change the flow of power among the various buses B 0   141  through BN  143 . In other words, controller  110  turns switches S 0   131  through SN  133  and switch SGND  134  ON and OFF in order to either allow energy from one or more of buses B 0   141  through BN  143  to be stored in reactive elements  120 , or to transfer stored energy in reactive elements  120  to any combination one buses B 0   141  through BN  143  or ground. The use of a controller such as described above enables bidirectionally power flow between two or more power components, and thus, is beneficial in circuits using a combined buck and boost topology, a flyback topology, or any other suitable circuit topology. 
   In one example, bus B 0   141  can be a DC wall adapter, bus B 1   142  can be a battery, and bus BN  143  can be a portable electronic device. It should be noted that although the use of a wall adapter results in external conversion of power from an AC power source to DC power, such conversion may be incorporated directly into the operation of a bidirectional power converter in accordance with the principles of the present invention. Accordingly, B 0   141  can be an AC power source, in which case some type of rectification would be included within the bidirectional power converter utilizing the AC power source. In the example above, controller  110  may be a current-mode controller that uses multiple control loops in order to OPEN and CLOSE the switches, and thereby control the transferring of power among the buses. As explained in greater detail below, controller  110  controls the flow of power among the various buses by adjusting the magnitude and direction of current flow through reactive elements  120 , from and to the various power buses in the circuit. 
   Additionally, it should be understood by those skilled in the art that not every switch shown in  FIG. 1  must necessarily be used. Generally, when certain switches are not needed, they may be replaced by shorts. For example, switch SGND in  FIG. 1  may be replaced with a short, depending on the arrangement of reactive elements  120 . 
     FIG. 2  shows a simplified bidirectional buck and boost converter in accordance with the principles of the present invention. Buck converters, also known as step-down converters, produce an output voltage that is smaller in magnitude than a supplied input voltage. On the other hand, boost converters, or step-up converters, provide an output voltage that is larger in magnitude than a supplied input voltage. 
   Circuit  200  includes controller  210 , buses B 0   241  and B 1   242 , switches S 1   231  and SGND  232 , and inductor  220 . The circuit shown in  FIG. 2  operates as a buck converter when passing power from bus B 1   242  to bus B 0   241 , left to right. Alternatively, it functions as a boost converter when passing power from bus B 0   241  to bus B 1   242 , right to left. Moreover, controller  210  controls switches S 1   231  and SGND  232 , and inductor  220  serves as the reactive element in which power may be stored and from which power may be taken. 
   In bidirectional converter  200 , because only a simple inductor  220  is used as the reactive element, the device may be fabricated with relatively small size and high density. Moreover, bidirectional converter  200  works well for bus voltages V 0 &lt;&lt;V 1  (V 0  is the voltage at bus B 0   241  and V 1  is the voltage at bus B 1   242 ). As V 0  approaches V 1 , however, the duty factor required for operating as a boost converter may become a problem. This problem may be alleviated, as explained, below, through the use of a flyback transformer instead of inductor  220 . 
   Similar to controller  110  of  FIG. 1 , controller  210  shown in  FIG. 2  controls the switching that causes power transfer in converter  200 . Additionally, the controller is once again a current-mode controller using multiple control loops in order to OPEN and CLOSE the switches, thereby controlling the direction of power flow. 
   The circuit shown in  FIG. 2  may also be expanded to include additional power buses.  FIG. 3  shows a bidirectional buck and boost converter with additional buses in accordance with the principles of the present invention. 
   For the circuit shown in  FIG. 3 , it should be noted that Isense bus  311  may convey one or more sense signals to controller  310  (as explained above for the circuit shown in  FIG. 1 ). It may be desirable, however, for Isense  311  to be a time multiplexed current sense signal which would be representative of the current that is individually supplied to any one of the power buses. This can be accomplished by providing for power transfer to or from only one of the power buses at any given time. Alternatively, separate sense signals can be used (one for each of the power buses) if power is being supplied from or delivered to more than one power bus at any given time. 
   In addition, persons of ordinary skill will appreciate that, assuming the normal voltage requirements for a buck or boost converter are met, bus B 0   341  shown in  FIG. 3  takes on special significance. This significance derives from the unvarying ability of bus B 0   341  to take power from or pass power to any of the other buses in the circuit. This is a result of bus B 0   341  being directly coupled to inductor  220 , and therefore, not having a switch controlling its connection to inductor  220 . Nevertheless, it remains possible to transfer power among the remaining buses. For example, bus B 1   342  can provide power to, or take power from, any combination of buses B 2   343  to BN  344 . Additionally, bus B 2   343  can provide power to, or take power from, bus B 1   342  in addition to any of buses B 3  (not shown) to BN  344 . However, when power is being transferred among buses B 1   342  to BN  344 , bus B 0   341  may absorb or provide additional power. For this reason, although the above mentioned and other bus power transfer combinations are possible, they should generally be avoided. 
   One manner in which the above power transfer combination problem can be avoided is to incorporate a switch between bus B 0   341  and inductor  220 . In this case, it is preferable to use a single-pole, double-throw switch. Furthermore, additional buses may be coupled to inductor  220  on the side coupled to bus B 0   341  without departing from the scope of the invention, and these additional buses also may or may not be separated from inductor  220  by switches. 
   It should be appreciated by those skilled in the art that the bidirectional buck-boost converter shown in  FIG. 2  and described above is a non-insulating converter. This type of converter is generally used in applications where voltage is only required to be stepped up or down by a relatively small amount, and in which it is acceptable for there not to exist electrical isolation between the input and output of the converter. In certain situations, however, it becomes desirable or even necessary to obtain isolation between the converter input and output. This isolation may be accomplished by removing the inductor and incorporating a transformer into the converter. Specifically, flyback converters are a version of buck-boost converters that incorporate transformer isolation while storing energy. 
     FIG. 4  shows a circuit diagram  400  of a bidirectional flyback converter in accordance with the principles of the present invention. Circuit  400  includes controller  410 , flyback transformer  420 , switches S 0   431  and S 1   432 , and buses B 0   441  and B 1   442 . 
   Although flyback transformer  420  of circuit  400  is represented using the same symbol as an ideal transformer, it is better described as a “two-winding inductor.” Flyback transformers, unlike ideal transformers, do not allow current to flow simultaneously in both windings. Instead, the magnetizing inductance of a flyback transformer assumes the role of the inductor of the buck-boost converter, and the magnetizing current is switched between the primary and secondary windings. 
   Furthermore, low-side N-channel MOSFET switching, or ground-side switching, simplifies the switch driver design in circuit  400 . In this case, the MOSFETs acting as switches have their respective gates connected to a controller, their drain terminals connected to the flyback transformer and their source terminals connected to ground via a sense circuit. 
   The main advantage of using N-channel MOSFETs results from the typically greater efficiency when compared to using a comparable on-resistance P-channel device (because less gate charge will be required for switching). Additionally, low-side switching with N-channel MOSFETs permits the use of MOSFETs with relatively low absolute maximum gate-source ratings, even in higher voltage applications. To further simplify the circuit design, common current sensing may be used to reduce the component count. For example, a sense resistor may be used, such as sense resistor  412  shown in  FIG. 4 . Alternatively, another suitable component may be used for this purpose. 
   An important consideration when dealing with converter  400  is the coupling coefficient of flyback transformer  420 . The windings of a flyback transformer are said to be coupled when the are arranged in such a manner that a changing magnetic field created by one of the windings can induce a current in the other winding. Moreover, the coupling coefficient of transformer  420  indicates the degree of coupling between the windings, and should be made as high as possible (i.e., the coupling coefficient should be as close to one as possible). 
   The coupling coefficient of transformer  420  is dependent on the portion of the total flux lines that cuts both the primary and secondary windings. Preferably, all the flux lines generated by the primary winding should cut the secondary winding, and vice versa. Lines of flux generated by one winding that do not link with the secondary winding are referred to as leakage flux, and result in having a voltage induced in the secondary winding that is less than would otherwise occur under ideal conditions. For this reason, efficiency is reduced as a result of leakage, and spikes may result at the drain of the switching device. 
   Bifilar winding can be used to reduce leakage inductance and improve the magnetic coupling of the windings, and therefore, is often used when it is important to achieve a superior coupling coefficient and to improve overall efficiency. In this situation, a simple integer turns ratio may be desirable. Nevertheless, the turns ratio of the windings of flyback transformer  420  can be adjusted, although the resulting duty factor should be taken into consideration. Furthermore, particularly in low voltage applications, consideration should be given to selecting a transformer with low winding resistance in order to improve efficiency at heavier loads. 
   It should be understood that the ratio of V 1  (the voltage at power bus B 1   442 ) to V 0  (the voltage at power bus B 0   441 ) in  FIG. 4  is independent on the turns ratio of the bifilar winding of flyback transformer  420 . Accordingly, the ratio of V 0  to V 1 , or any other combination of power buses shown in the figures described below, can be manipulated regardless of the turns ratio of flyback transformer  420 . 
   A particular benefit of the bidirectional converter shown in  FIG. 4  over the converters shown in  FIGS. 2 and 3  is that converter  400  will maintain normal operation regardless of whether bus voltage V 0  is greater or less than bus voltage V 1 . This can be important when either V 0  or V 1  originates from a battery, because some batteries (for example Lithium-Ion batteries) show large cell voltage variation as they discharge. 
   A similar flyback converter topology to that shown in  FIG. 4  may also be used for applications involving the transfer of power among more than two components. Persons of ordinary skill in the art will appreciate that additional buses may be added to the converter of  FIG. 4 , as shown in  FIG. 5 . Circuit  500  is a circuit diagram of a bidirectional flyback converter with additional buses in accordance with the principles of the present invention. Circuit  500  includes controller  510 , transformer  520 , switches S 0   531 , S 1   532 , S 2   533  and SN  534 , and buses B 0   541 , B 1   542 , B 2   543  and BN  544 . 
   Additional DC connections are made to flyback transformer  520  by adding multiple secondary windings. As shown, on either side of each winding, a switch and its corresponding bus voltage is connected. A person skilled in the art will appreciate that the polarity dots on the transformer are correct in situations where the voltages on all buses have the same polarity, for example, V 0 -VN are all positive or V 0 -VN are all negative. Moreover, the preferred power transfer. combinations involve power transfer between bus B 0   541  and the remainder of the buses, but in some instances other combinations are also acceptable. Once again, as with the circuits described above, Isense bus  511  may convey one or more sense signals to controller  510 . Assuming that only one power bus is receiving power from or supplying power to flyback transformer  520 , a single current sense signal can be used (where the sense signal is sampled at certain time intervals). Alternatively, multiple sense signals can be used when more than one power bus is receiving or supplying power at any given time. 
   It should be understood by those skilled in the art that each possible power flow combination of the aforementioned bidirectional power converters may have a unique application, unique power converter topology, unique demands on transient response and unique input and output voltages or currents. For example, a bidirectional flyback converter may be operating in one of two different modes, namely, discontinuous conduction mode or continuous conduction mode. In the former, all energy stored in the first winding during the ON (energy storing) time is completely delivered to the other winding and to the load before the next cycle, and there may also be dead time between the instant the secondary current reaches zero and the start of the next cycle. A circuit operating in discontinuous mode is typically designed to enter into continuous mode as the output current level is increased beyond a predetermined value. When continuous mode is entered into, there remains energy in the secondary winding at the beginning of the next cycle. 
   Furthermore, because of the many different requirements for different power flow combinations, each power transfer combination may require different feedback signals and compensation (examples of which are provided below), except that a sense current is typically required in order to determine the direction of current flow.  FIG. 6  is a simplified circuit diagram of an error amplifier using a shared-amplifier arrangement with sense signal and compensation selection that may be used in a control circuit, as described below, in accordance with the principles of the present. In the examples provided below, current-mode controllers are used in determining the direction of power flow among the various power buses. The present invention is not limited in this respect, however, as voltage-mode controllers (or any other suitable type of controller or combination of controllers) that can use signals representative of average current flow may also be used to control the flow of power among various power buses in a bidirectional power converter. 
   Depending on the desired power flow combination (i.e., the source(s) of power and the destination(s) of the power being transferred), decoder  610  selects the sense signals and compensation (feedback network) needed. In the embodiments discussed below, a direction signal serves to select the appropriate sense signal and feedback network. Moreover, in accordance with the principles of the present invention, and as shown in  FIG. 6 , there may be up to M voltages or voltage representations of sensed currents in the power converter (shown as sense 1   631  through senseM  633 ), in addition to up to P different feedback networks. It should be understood by those skilled in the art that any combination of sense signals may be used with any combination of feedback networks. Moreover, the complexity of this circuit can be traded off against the transient performance. 
   As shown in  FIG. 6 , switch S 1   662  couples one of the sense signals to the inverting input of error amplifier  620  (depending on the selection by decoder  610 ). Decoder  610  also selects the appropriate feedback network by causing switch S 0   661  to close a particular feedback loop (i.e., one of feedback networks  651  through  653 ). Accordingly, based on circuit conditions, decoder  610  control switches S 1   662  and S 0   661  in order to produce an error signal (to be used in a manner described below). 
     FIG. 7  shows an embodiment of a portion of a controller, or control circuit, to be used in the control circuits for  FIGS. 8-9  described below. The circuit shown in  FIG. 7  uses the principles explained above for  FIG. 6 , and may be used by a bidirectional power converter having particular sense signals and feedback networks in place (as shown). Optional current amplifier  720  takes the voltage potential across a current sense resistor and adjusts and (potentially). creates an offset signal as required. 
   When a converter using the circuit shown in  FIG. 7 , such as those shown in  FIGS. 8-9  and described below, is operating as a charger, switch  781  is controlled in such as manner that error amplifier  740  uses the signal provided by amplifier  720  (which is supplied through an input network comprising resistor  763 ) and uses a feedback network comprising capacitor C 1   771 . In this manner, when the circuit of  FIG. 7  is being used by a converter that is operating as a charger, error amplifier  740  acts as an integrator with the signal and feedback network described above. 
   Alternatively, when the converter using circuit  700  is operating as a regulator, error amplifier  740  uses the signal provided by the voltage sense line (supplied through an input network comprising a voltage divider made up of resistors R 4   764  and R 5   765 ) and uses a feedback network comprising resistor R 6   766  and capacitor C 2   772 . In this manner, when circuit  700  is being used by a converter operating as a regulator, error amplifier  740  acts as an integrator with zero with the signal and feedback network described above. Using an integrator with zero configuration when the converter is operating as a regulator, as opposed to a simple integrator configuration as in the case above when the converter is operating as a charger, provides a faster transient response. Although having a faster transient response is particularly important when the converter is operating as a regulator as opposed to a battery charger, the integrator with zero configuration (or another suitable configuration) may be used in all modes of operation of the converter. 
   Moreover, for purposes of simplification, it is assumed that in circuit  700 , error amplifier  740  can drive both feedback networks (i.e., the integrator and integrator with zero feedback networks). Thus, an output selection switch is not necessary as shown in  FIG. 6  (see component S 0   661  and the description provided above), and only input switch S 1   781  is used. The present invention is not limited in this respect, and it should be understood by those skilled in the art that an output selection switch (to select an appropriate feedback network) may be implemented with minimal effort. 
   Oscillator  710 , current amplifier  720 , pulse-width modulation (PWM) comparator  730  (which receives a current ramp from oscillator  710 ) and PWM latch  750  provide the remaining components for circuit  700 . A driver, not shown in  FIG. 7 , maintains a connection from circuit  700  and uses a zero current comparator for synchronous rectifier operation and logic to route and buffer the output of the PWM latch to the switches. The driver may also include anti-cross-conduction circuitry, logic inversions, and other logic functions. 
     FIG. 8  shows a circuit diagram of a simplified current-mode buck battery charger and boost regulator power circuit used to power an electronic device in accordance with the principles of the present invention. 
   The circuit of  FIG. 8  uses control circuit  801  (comprising the components of circuit  700 ) to control the transfer of power among the various power buses (i.e., wall adapter  810 , electronic device  830  and battery  820 ). When wall adapter  810  is providing ample power and maintains a connection with the circuit, it provides electric device  830  with power, and rechargeable battery  820  is charged by the buck charger comprising inductor  890 , switch S 1   891  and switch SGND  892 . (capacitors C 1   881  and C 2   882  are included for filtering and to reduce electromagnetic interference). If wall adapter  810  is either disconnected from the circuit, not supplying adequate power, or turned OFF (i.e. not supplying any power), power is taken from battery  820  and supplied to electronic device  830  by the boost converter formed by the same power devices as before (switches S 1   891  and SGND  892 , along with inductor  890 ). The interconnection of components is not altered when the circuit changes its operation between buck and boost modes, but the power flow reverses direction because the error signal from RSENSE  851  causes control circuit  801  to operate switches S 1   891  and SGND  892  in such a manner that enables this to happen. 
   Control circuit  801  operates in the following manner. If the circuit in  FIG. 8  is operating as a battery charger, control circuit  801  measures the average voltage drop across RSENSE  851  during the battery charging phase of the power converter in order to control the charging of battery  820 . On the other hand, if circuit  800  is operating as a regulator (power taken from battery  820  and supplied to electronic device  830 ), control circuit  801  uses the voltage divider comprising resistors  852  and  853  to control the regulation of the voltage being supplied to electronic device  830 . 
   The direction signal emerging from direction comparator  845 , which measures the ability of wall adapter  810  to supply power, is fed into driver  860 . Additionally, the direction signal controls switch SW 1   893 , thereby selecting the input network and feedback network to be used by error amplifier  843  in receiving the sense signal. As explained above, an separate output switch is not necessary to select the appropriate feedback network, but may easily be implemented. Moreover, for battery charging, SW 1   893  preferably ensures that error amplifier  843  acts as an integrator (i.e., the feedback network includes a capacitor), while during voltage regulation it preferably ensures that error amplifier  843  acts as an integrator with zero (i.e., the feedback network includes a resistor in series with a capacitor), resulting in a faster transient response. 
   At the beginning of each oscillator cycle, PWM latch  872  is set. Depending on the desired direction of power flow, the appropriate switch is turned on. The remaining components that make up control circuit  801  for this converter ensure that the current flowing through inductor  890  remains below a predetermined level while power is being transferred among the circuit components. 
   A consideration relating to compensation in converter  800  is that the poles and zeros of the power circuit are different for the circuit when operated as a buck converter as opposed to a boost converter, the result being that the circuit shown in  FIG. 8  has a different power circuit transfer function when operating in different modes. For example, there is no right half plane zero in the circuit transfer function when converter  800  is operating as a buck converter in continuous conduction mode, but there is a right half plane zero in the circuit transfer function when it is operating as a boost converter. This change should therefore be taken into consideration in order to enable optimal performance of converter  800  depending on the application, for example, by changing the location of the poles and zeros of the circuit transfer function. Changing the poles and zeros may be implemented by customizing the various circuit components (e.g., changing the components associated with error amplifier  843 ). Accordingly, consideration can be given to the fact that converter  800  has the potential for considerably better transient response when operating as a buck converter as opposed to operating as a boost converter. Moreover, load and line transient response, as previously mentioned, typically needs to be much quicker when circuit  800  is operating as a regulator rather than when operating as a battery charger. 
     FIG. 9  shows another embodiment of a simplified current-mode buck battery charger and boost regulator power circuit used to power an electronic device. The bidirectional power converter shown in  FIG. 9  is substantially identical to the one in  FIG. 8 , except that direction comparator  845  is replaced with direction comparator  945 , and a diode  996  is added for the purpose of allowing direction comparator  945  to detect whether or not wall adapter  810  is supplying power to the circuitry. Accordingly, instead of taking a direct measurement of the power being supplied (or lack thereof) from wall adapter  810  as in the circuit of  FIG. 8 , direction comparator  945  in  FIG. 9  provides a signal indicative of the voltage drop across diode D 1   996  for the purpose of determining the available power from wall adapter  810 . In this manner, the power source is chosen (either wall adapter  810  or battery B 1   820 ) in order to deliver the necessary power to the remaining components. 
     FIG. 10  shows a circuit diagram of a simplified bidirectional flyback converter for positive supply regulation and battery charging in accordance with the principles of the present invention. As in the circuits shown in  FIGS. 8-9 , converter  1000  includes a control circuit  1001  that uses a current feedback signal. Moreover, the embodiment shown in  FIG. 10  works especially well for the latest generation of MOSFETs that have low absolute maximum gate-source voltages (usually less than the maximum drain source voltages). Additionally, gate-source voltage limiting may be accomplished via ground-referenced regulation of the gate drive supply voltage. 
   In circuit  1000 , voltage selection network  1064  selects the power source as either battery B 1   1020  or wall adapter  1010  based on the voltages available and creates an output voltage for all the remaining devices. One typical scheme uses the voltage of the two available voltages that is greater in magnitude to create voltage VMAX  1067 . Another technique would be to take the lower of the two available voltages from battery B 1   1020  and wall adapter  1010 , as long as it exceeds the minimum voltage needed by the remaining devices. This technique may provide added efficiency but also increases the complexity of voltage selection network  1064 . 
   VDRIVE  1072  is an optional regulated voltage linearly regulated down from VMAX  1076  which may be needed when power MOSFETs M 1   1062  and M 2   1063  have low absolute maximum gate-source voltages. In low voltage applications, Schottky diodes  1068  and  1069  are placed in parallel (as shown in  FIG. 10 ) with the drain and source of the respective MOSFETs in order to prevent the body diode from turning on. This configuration preferably improves efficiency by helping to eliminate losses due to reverse recovery in these body diodes and forward voltage drops during the break interval. Additionally, snubbers SN 1   1065  and SN 2   1066 , which are generally series RC (i.e., resistive-conductive) snubbers, keep the drain-source voltages of transistors M 1   1062  and M 2   1063  within acceptable limits. Moreover, bifilar winding (as previously described) in flyback transformer  1061  may be used to reduce the need for aggressive snubber design. 
   It should be understood by those skilled in the art that a regulated wall adapter may be used in place of unregulated wall adapter  1010 , thereby eliminating the necessity for low dropout regulator  1084 . In that case, the output to the circuitry and other devices (e.g., a portable electronic device) could be taken. directly across capacitor C 2   1051  shown in  FIG. 10 . The invention is not limited in this manner. 
   During the operation of power converter  1000 , the direction signal emerging from comparator  1031  determines whether converter  1000  operates as a battery charger or voltage regulator. Based on this direction signal, switch SW 2   1092  selects the appropriate sense signal, input network, and feedback network for error amplifier  1034  for either battery charge current control, or regulator output voltage control. Once again, an output switch is not used to select the appropriate feedback signal, but the present invention is not limited in this respect. The direction signal originating from comparator  1031  also feeds driver  1070  and determines whether M 1   1062  charges (stores energy in) transformer T 1   1061  and M 2   1063  discharges T 1   1061 , or whether these roles for M 1   1062  and M 2   1063  are reversed. 
   During battery charging, switch SW 2   1092  is in the down position, and resistors R 7   1047  and R 8   1046 , along with capacitor C 4   1054 , create a simple integrator with error amplifier  1034  and also set up nominal charge current. At the start of the oscillator cycle, PWM latch  1082  is set, causing M 2   1063  to turn on. When the switch current reaches a predetermined level, as set by the control voltage at the output of error amplifier  1034 , M 2   1063  turns off and M 1   1062  turns on. Additionally, when transistor M 2   1063  is off and a battery charging current is flowing through resistor R 4   1044 , switch SW 1   1099  samples the current sense signal. In this case, battery charging current can flow either through the body diode or from drain to source of transistor M 1   1062 . For this reason, RDRIVE is the signal to use for observing the charging current. 
   When circuit  1000  is being used for voltage regulation (i.e., power is being supplied by the battery), switch SW 2   1092  is in the up position and resistor R 6   1048  and capacitor C 3   1053  create an integrator with zero with error amplifier  1034 , while resistors R 1   1042  and R 2   1043  set up the nominal output voltage of the regulator to be compared with voltage reference level  1083 . 
   During voltage regulation, M 1   1062  turns on at the start of each oscillator cycle. Moreover, when the transformer current reaches a predetermined level as set by the output of error amplifier  1034 , M 1   1062  turns off and M 2   1063  turns on. Current amplifier  1033  increases the amplitude of the current sense signal (observed as a voltage drop across resistor R 4   1044 ) while rejecting noise arising from M 1   1062  and M 2   1063  gate charge, switching transients from primary and secondary currents of transformer T 1   1061 , common-mode noise potentially from other circuits, and ringing that may result from imperfect coupling between transformer windings. As shown in  FIG. 10 , the common mode voltage level is supplied by voltage reference level  1083 , and the two output terminals of current amplifier  1033  which connect to resistors R 5   1045  and R 7   1047  provide complementary signals. Additionally, amplifier U 6   1033  may also include blanking synchronized to the gate drive signals and/or bandwidth limiting. 
   When transformer T 1   1061  has fully dumped its energy, the MOSFET that was serving as the synchronous rectifier may be shut OFF. This is accomplished through the use of zero current comparator  1032 , which shuts OFF either M 1   1062  or M 2   1063  at the appropriate time. This is commonly done when power converter  1000  is expected to enter discontinuous conduction at light loading in order to improve the efficiency at lighter loads. 
   Comparator  1035  serves as a PWM comparator for peak-current mode control power converter  1000 . Oscillator  1081  triggers the start of a power conversion cycle by setting PWM latch  1082  (driving Q high), assuming the reset-dominant input of PWM latch  1082  is not high. This causes the MOSFET (either M 1   1062  or M 2   1063  depending on the direction of power transfer) to turn on, thereby charging transformer T 1   1061 . As the energy stored in transformer T 1   1061  increases, the current through sense resistor R 4   1044  also increases, and, therefore, the voltage at the non-inverting input of PWM comparator  1035  also increases. When the voltage at the non-inverting input of PWM comparator  1035  exceeds the voltage on the inverting input, the output of PWM comparator  1035  goes high, resetting PWM latch  1082 . As a result, the MOSFET that was charging transformer T 1   1061  is turned off. In addition, it should be noted that oscillator  1081  provides PWM comparator  1035  a ramp current (similar to oscillator  710  described above). However, a voltage ramp could also be used with minor circuit modifications. The same is true for each of the bidirectional power converter circuits described in accordance with the principles of the present invention. 
   At this point, neither of MOSFETs M 1   1062  or M 2   1063  are turned on, and transformer T 1   1061  begins to discharge through the body diode of the opposite MOSFET. This is a result of the flyback action and polarity of the windings of the transformer. After the break time (i.e., the time that MOSFETs M 1   1062  and M 2   1063  are off) has elapsed, the opposite MOSFET turns on connecting transformer T 1   1061  to the load side of the circuit. If at any time during this part of the power conversion cycle the inductor current reaches zero, zero current comparator  1032  shuts off the MOSFET that was connecting transformer T 1   1061  to the load side of the circuit. The circuit then waits for oscillator  1081  to provide a pulse to the S input of PWM latch  1082 . When this pulse is provided to the S input of PWM latch  1082 , another complete power conversion cycle begins. 
   Capacitors C 1   1052  and C 2   1051  are included at least for the purpose of filtering the current pulses. from the power converter, reducing electromagnetic interference, ripple voltage, and ripple current. Moreover, the sizing of capacitors C 1   1052  and C 2   1051 , which preferably have low effective series resistance and inductance, should be selected depending on desired circuit characteristics (e.g., ripple voltage amount). 
   It should be noted that transformer saturation, which typically can occur at high output currents and extreme duty cycles, should be avoided under all operating conditions and combinations. In addition, although not shown, duty cycle limiting and pulse skipping at light loading is generally recommended for this topology. For example, Burst Mode™ of operation (hereinafter, “Burst Mode”) may be entered into in order to provide higher operating efficiency. Examples of regulators that use Burst. Mode are Linear Technology Corp.&#39;s 1435 and 1735 series products. 
   When operating in Burst Mode, the power MOSFETs operate intermittently at light loads, thereby increasing efficiency by minimizing switching losses. A Burst comparator may be used to determine when Burst Mode should be enabled, during which time the MOSFETs are not switched according to their normal cycles. It also determines when Burst Mode should be disabled, at which time normal circuit operation is resumed. Due to the introduction of broadband noise during Burst Mode operation, in particular during battery charging, it may be desirable to use normal operation. 
     FIG. 11  shows a circuit diagram of another embodiment of a simplified bidirectional flyback converter for positive supply regulation and battery charging in accordance with the principles of the present invention. The bidirectional power converter shown in  FIG. 11  is substantially identical to the converter of  FIG. 10 , except that direction comparator  1031  is replaced with direction comparator  1131 , and a diode  1185  is added for the purpose of allowing direction comparator  1031  to detect whether or not wall adapter  1010  is supplying power to the circuitry. 
   In bidirectional converter  1100 , direction comparator  1131  determines if a wall adapter is connected and operating (i.e., supplying sufficient power) by sensing the voltage drop across diode D 1   1185 . This requires the voltage supplied by wall adapter  1010  to be several hundred millivolts higher (at room temperature) than the voltage the bidirectional converter supplies in order to adequately forward bias diode. D 1   1185 . This reduces net efficiency (wall adapter to battery) slightly. 
   Smaller differential voltages (wall adapter to circuit) may be used to sense a powered wall adapter by using resistive wall adapter detection. This is accomplished by replacing diode D 1   1185  with a current sense resistor (not shown). 
   There are, however, several disadvantages to using resistive wall adapter detection. First, the drop across the current sense resistor would vary with the current supplied by wall adapter  1010 . Although in a current-limited battery charger this may not be a problem until the float voltage is reached, the drop in the sense resistor nonetheless should be carefully chosen. Second, offset in wall adapter detection comparator  1031  would become more critical, as would the common mode rejection ratio, filtering, and circuit layout (Kelvin connections to the current sense resistor, etc.) Third, R 3   1041  would load the power converter when power is being supplied by battery  1020 , and thus, resistor R 3   1041  would have to be made relatively large in order for it not to impact the efficiency of the converter. One possible solution to this problem is to replace R 3   1041  with R 1   1042  and R 2   1043  by moving R 1   1042  and R 2   1043  to the right of the current sense resistor (and eliminating R 3   1041 ). Alternatively, the diode-based wall adapter detection circuit shown in  FIG. 11 , combined with an inexpensive unregulated wall adapter  1010 , provides a simple and cost-effective design. 
     FIG. 12  shows a circuit diagram of a particular embodiment of driver circuitry that may be used in accordance with the principles of the present invention. Driver circuitry  1200  may be used, for example, in bidirectional power converters such as shown in  FIGS. 8-11  and described above. 
   The one embodiment of driver circuitry, like driver circuitry  1200 , corresponds to component  860  in  FIGS. 8-9  and component  1070  in  FIGS. 10-11 . Driver circuitry  1200  includes a plurality of NOR gates (components  1231 - 1234  and  1236 - 1237 ), XOR gate  1235 , level shifter circuits  1241 - 1243 , buffers  1251 - 1252  and inverters  1221  and  1222  interconnected as shown in  FIG. 12 . Driver circuitry  1200  provides the regulator side and battery side switches of the regulators shown in  FIGS. 8-11  with controlling signals RDRIVE  1261  and BDRIVE  1262 , respectively, which are the signals that control the direction of power flow in the bidirectional power converters. 
   Identical circuits  1251  and  1252  in  FIG. 1200 , along with NOR gates  1236  and  1237 , provide two levels of anti-cross conduction circuitry. These circuits are used to ensure that the controlling signals, RDRIVE  1261  and BDRIVE  1262 , do not allow the external power switches which they are controlling (e.g., switches  1068  and  1069  in  FIG. 10 ) to be ON at the same time. Accordingly, when either the regulator side switches or the battery side switches of the bidirectional converters described above are made to turn ON, the other switch is made to turn OFF. 
   For each of circuits  1251  and  1252 , the components therein are configured such that the two internal MOSFETs are precluded from being simultaneously ON. This is accomplished in circuit  1251  through the use of feedback signals  1271  and  1272 , which respectively drive NAND gates  1282  and  1281 . Similarly, in circuit  1252 , feedback signals  1273  and  1274  drive NAND gates  1284  and  1283 , respectively. Accordingly, in circuit  1251 , N-channel MOSFET  1254  does not turn ON before P-channel MOSFET  1255  has turned OFF, and vice versa. The same situation is present for internal MOSFETs  1255  and  1256  of circuit  1252 . Moreover, feedback signals  1275  and  1276 , which drive NOR gates  1237  and  1236 , respectively, are. responsible for preventing RDRIVE  1261  and BDRIVE  1262  from turning the external power MOSFETs ON at the same time. 
   Level shifter circuits  1241 - 1243  shown in  FIG. 1200  are cross-coupled level shifters that are used to translate inter-logic signals at a lower voltage level to a higher voltage level (as necessary to drive the regulator and battery-side MOSFETs). The purpose of these cross-coupled level shifters is to keep the voltages low until it is required that the voltages be raised in order to drive the regulator and battery-side MOSFETS. 
   Moreover, the circuit shown in  FIG. 12  has four significant inputs (not shown in previous figures) that allow driver  1200  to provide signals RDRIVE  1261  and BDRIVE  1262  in order to control the switches of the bidirectional power converter as desired. Depending on circuit conditions, these inputs are used by driver circuitry  1200  to control the regulator and battery-side MOSFETs. Accordingly, these four inputs control whether power is transferred either from right to left (when the converter is acting as a battery charger) or left to right (when the converter is acting as a DC-DC converter) through the reactive elements. 
   Store signal  1214  is used to determine when energy is stored in the reactive elements (i.e., when power from any of the available power buses capable of supplying power will be used to store energy in the windings of the transformer or in the inductor acting as the reactive element in the bidirectional power converter). In other words, when store signal  1214  is a logic high, this input triggers the start of a new power conversion cycle, and power is then transferred to the reactive elements to be stored. When store signal  1214  is a logic low, however, the energy previously stored in the reactive elements is transferred to any combination of the available power buses. 
   In the converters of  FIGS. 8-9  and  FIGS. 10-11 , store signal  982  originates from PWM latch  872  and PWM latch  982 , respectively. In driver  1200 , store signal  1214  feeds XOR gate  1235  and determines which of NOR gates  1236  and  1237  is a logic high or a logic low. In this manner, store signal  1214  causes the MOSFETs of the bidirectional power converter to turn ON or OFF, thereby determining which side of the converter is supplying power to the reactive elements. 
   Synchronous rectification, or rectifier input  1212 , on the other hand, receives a signal from a zero current comparator (component  841  in circuits  800  and  900  and component  1032  in circuits  1000  and  1100 ). This signal allows the driver to facilitate rectification using the switches within the respective circuits, thereby providing control over the power transfer. 
   In order to control the source of power for charging the reactive elements (e.g., either the battery or the wall adapter for the case of the circuits described above), driver circuitry  1200  also includes a selection signal input. Selection signal  1211  indicates whether or not a wall adapter is supplying power by sensing the presence or absence of a voltage drop across a diode. This signal is provided to driver circuitry  1200 , and also selects the sense signal for the error amplifier found in the bidirectional power converter circuitry. For example, in  FIG. 8 , direction comparator  845  provides driver  860  and switch SW 1   893  with a signal depending on the available power from wall adapter  810 . Similarly, in  FIG. 10 , comparator  1031  provides driver  1070  and switch SW 2   1092  with a signal depending on the available power from wall adapter  1010 . Moreover, selection signal  1211  may also be used in determining that certain operating conditions are met (e.g., that overcharging of a power bus is not occurring). If any of the predetermined operating conditions are not met, selection signal  1211  may stop the transfer of power among the various power buses. 
   Finally, in the embodiment shown in  FIG. 12 , driver circuitry  1200  has as an input a burst mode signal input  1213 . As described earlier, Burst Mode operation may be incorporated into the embodiments of the bidirectional power converters presented according to the principles of the present invention. Accordingly, when the Burst Mode signal is asserted (i.e., the Burst Mode signal is a logic high), this signal would indicate to the driver that the converter circuitry should enter into sleep mode, at which time both external power MOSFETs are off, thereby improving efficiency. 
   Further consideration when dealing with circuits that use current-mode controlling (e.g., applicants&#39; claimed bidirectional power converter) should preferably be given to the use of slope compensation. In particular, when dealing with the switching in a fixed frequency, peak-current mode, pulse width modulated power supply, open loop instability or subharmonic oscillation can occur when the duty cycle (i.e., the ratio of the switch pulse width to the oscillator period) exceeds about 50%. 
   A typical solution to this problem is to use a slope compensation signal, derived from the oscillator, to ensure stability regardless of duty cycle. The slope compensation signal can be applied, for example, by either adding or subtracting a portion of the oscillator signal to the current sense signal. Moreover, this can be done by either adding or subtracting a portion of the oscillator signal to the non-inverting terminal of the PWM comparator or the inverting terminal of the PWM comparator. 
   The slope compensation signal described above causes the perceived rate of current rise in the reactive element to be greater than the rate of current fall, thereby allowing the regulator to operate at duty cycles greater than 50% without becoming unstable. In one embodiment, a slope compensation signal proportional to the duty cycle may be added to from the sense signal to increase the rate of current rise perceived by the control circuitry. As the slope compensation signal progresses towards its peak, the flow of current through the switch is impeded and an increase in the rate of current rise in the reactive element is perceived, allowing the converter to operate at duty cycles greater than 50% without losing stability. 
   Moreover, the reactive element current down slope (i.e., the current when the reactive element current is decreasing) is a function of the output voltage, or the output voltage minus the input voltage (depending on the power converter topology). For this reason, when the power converter voltages vary widely, the typical approach has been to add enough slope compensation to prevent instability even under the worst case operating conditions. Therefore, the result is excessive slope compensation being used under more normal operating conditions. This is particularly undesirable when dealing with bidirectional power converters, in which power converter voltages commonly have a large range of duty factors. Moreover, because the actual maximum current that can pass through the output switch is generally a function of the amount of slope compensation being used, the problem of overcompensation becomes especially detrimental in bidirectional power converters and causes the actual maximum current that can pass through the output switch to decrease proportionally as the duty cycle increases. 
     FIG. 13  is a circuit diagram of an adaptive slope compensation network, in accordance with the principles of the present invention, that is capable of substantially eliminating the effects of overcompensating as described above.  FIG. 13  accomplishes this by using linear slope compensation waveforms (i.e., adjusting the amplitude of the ramp signal being used by using a linear waveform and adjusting the gain as necessary), and thereby provides an effective slope compensation signal that varies depending on the duty cycle. 
   In particular, the slope compensation network shown in  FIG. 13  has a slope compensating ramp that is amplitude controlled so that it is always optimal. It should be understood that although linear slope compensation waveforms are used in the circuit of  FIG. 13 , a variable slope compensation signal can also be created by using non-linear waveforms. Nevertheless, when dealing with an extremely wide range of voltages, it is often beneficial to use linear waveforms to avoid excessive errors that can result due to approximation errors associated with non-linear waveforms. 
   Circuit  1300  shown in  FIG. 13  includes a multiplier core of four transistors (components  1341  through  1344 ), bias currents Iramp  1381 , Igain  1382  and Ibias  1383 , and a fixed bias V DD    1371  interconnected as shown. In circuit  1300 , Vbe 4 =Vbe 3 +Vbe 1 −Vbe 2  (where Vbe 4  is the voltage differential between the voltages at the base and the emitter of transistor  1344 , etc.). Ignoring the base currents (which are negligible), assuming well matched transistors and ignoring other effects, the above equation produces the following equation: V t *ln(Iout/Is)=V t *ln(Iramp/Is)+V t *ln(Igain/Is)−V t *ln(Ibias/Is). After simplifying this equation, we are left with Iout=(Iramp*Igain)/Ibias. Accordingly, controlling the various bias currents allows provides adaptive slope compensation. 
     FIG. 14  shows a circuit diagram of a particular embodiment of an adaptive slope compensation network for a bidirectional power converter in accordance with the principles of the present invention. The multiplier core transistors  1341  through  1344  are labeled and configured in the same manner as in circuit  1300 , and selection signal  1211  and burst mode signal  1213  are the same signals as described above for  FIG. 12 . C 1  in circuit  1400  provides compensation of the loop formed by transistor Q 1   1341  and Q 3   1343 . Resistors R 1   1421 , R 2   1422  and R 3   1423 , and transistors Q 5   1445  and Q 6   1446  serve as a voltage to current converter with having first order voltage/current non-linearity compensation (caused by Q 5   1445 ). 
   Assuming conduct signal  1474  is a logic high, MOSFETs M 1   1451  and M 2   1452  select the correct gain control signal from either V BAT  or V REG  for the slope compensation (based on what voltage is determining the reactive elements downslope). These MOSFETs are turned ON and OFF, respectively, by NOR gates  1431  and  1432  (which are controlled by selection signal  1211 ). Furthermore, logic inverters  1411  and  1412  provides decoding for voltage selection and shutdown, and all remaining transistors and mosfets (labeled components  1453 - 1456 , and  1461 - 1462  in  FIG. 14 ) provide either current mirroring or shutdown in circuit  1400 . 
   Circuit  1400 , using the components described above and interconnected as shown in  FIG. 1400 , provides slope compensation waveform  1475  which is used in the bidirectional power converter to ensure stability, regardless of duty cycle, and allows this to be done without resulting in overcompensation. Accordingly, waveform  1475  can be either supplied to the non-inverting terminal or the inverting terminal of the PWM comparator (e.g., component  1035  in  FIG. 10 ), depending on whether it is preferable to add a portion of the oscillator signal to the current sense signal or rather, to subtract a portion of the oscillator signal. 
   The above described embodiments of the present invention are presented for purposes of illustration and not of limitation, and the present invention is limited only by the claims which follow.