Abstract:
In simultaneous reception of overlapping signals sharing a single spreading sequence, packet start times are found by separately demodulating, low pass filtering and despreading in-phase and quadrature components. Despread in-phase and quadrature signals are provided to sychronization filters connected in parallel and arranged in two identical pairs. Outputs from filters in different pairs are added, and the sums are squared. The squared signals are added. An exceeded threshold of the output from the last adder delineates a packet start. Knowing packet start times facilitates separating the overlapping inbound signals.

Description:
This application claims the benefit of U.S. Provisional Application No. 60/082,863, filed Apr. 24, 1998. 
    
    
     BACKGROUND OF THE INVENTION 
     SUMMARY OF THE INVENTION 
     In a Spread ALOHA receiver, when the received signal is a linear combination of the transmitted signals the detection of multiple overlapping packets is simplified. In the present invention, equations are derived that describe a QAM modulation and demodulation process in order to specify a common notation to be used in the network design. In the case of a single transmitter the derived equations are well known. For the case of a Spread ALOHA multiple access channel however, interest lies in the simultaneous reception of many overlapping signals. The problem does not appear to have been treated in the literature. 
     A receiver for spread bit packet signal transmissions has an input line with a QAM modulator connected thereto. The QAM modulator has a splitter connected to the input line for separating the input into first and second paired inputs. First and second multipliers are connected to the first and second pair inputs. A numerically controlled oscillator is connected to the first multiplier. A 90° phase shifter is connected to the numerically controlled oscillator and is connected to the second multiplier. First and second output lines are connected to the first and second multipliers. First and second low pass filters are connected to the first and second output lines for removing high frequencies from the outputs. First and second demodulated signal lines are connected to outputs of the first and second low pass filters, and first and second despreading filters are connected to the first and second demodulated signal lines. First and second despread signal lines are connected to outputs of the first and second despreading filters. First and second pairs of synchronization matched filters are connected to the first and second despread signal lines for producing packet synchronization sequence outputs. 
     A preferred method of information detection in a single spreading sequence receiver comprises receiving packet and chip signals in QAM modulated carrier wave signals with in-line and quadrature components. The received signals are supplied to first and second multipliers. Output is supplied from a numerically controlled oscillator to the first multiplier and to a 90° phase shifter. An output is supplied from the 90° phase shifter to the second multiplier. Output signals are supplied from the first and second multipliers to first and second low pass filters, and high frequency signals are removed in the low pass filters. First and second outputs are provided from the first and second low pass filters to first and second despreaders. First and second despread signals are provided from output of the first and second despreaders to inputs of first and second pairs of synchronization filters. 
     In one preferred embodiment, inputs of first and second synchronization filters are connected in parallel and the first despread signal is supplied from the first despreader to the inputs of the first and second synchronization filters. Second and third synchronization filters are connected in parallel to an output of the second despreader. Despread signals are supplied from the output of the second despreader to inputs of the third and fourth synchronization filters. Outputs of the first and third synchronization filters are supplied to a first adder. Outputs of the second and fourth sychronization filters are supplied to a second adder. An output of the first adder and an output of the second adder are squared, and squared outputs from the first and second squarers are added in a third adder. A threshold of an output from the third adder is detected as a start of a packet. 
     These and further and other objects and features of the invention are apparent in the disclosure, which includes the above and ongoing written specification, with the claims and the drawings. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic representation of a QAM modulator. 
     FIG. 2 is a schematic representation of a QAM demodulator. 
     FIG. 3 is a schematic representation of a multiple access QAM demodulator. 
     FIG. 4 is a schematic representation of synchronization matched filters. 
     FIG. 5 is a schematic representation of a multiple access QAM demodulator and packet detector. 
     FIG. 6 is a schematic representation of a multiple access QAM demodulator and packet detector. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     Transmitter for One Signal 
     Assume we have two bandlimited signals, a I (t) and a Q (t) in transmitter  40 . The QAM modulation  44  of these signals  41 ,  51  is accomplished by multiplying  43 ,  53  the signals  41 ,  41  by the output  45  of a numerically controlled oscillator (NCO)  47  while the quadrature multiplication  53  is shifted 90 degrees in phase. A block diagram of a QAM modulator  44  is shown in FIG.  1 . 
     The output  55  of the phase shifter  57  is used in multiplier  53 . The outputs  49  and  59  of the multipliers  43  and  53  are added  62  to produce an output signal  61 . 
     In the usual situation, the in-phase  41  and quadrature  51  signals, a I (t) and a Q (t), are bandlimited to a frequency much less than the carrier frequency ω 0 . The phase shift, θ 0 , of the NCO  47  is assumed to be a random variable and not a function of time. 
     Receiver for One Signal 
     The receiver  60  has a demodulator  64  for signal  61 , as shown in FIG.  2 . The overall structure of the demodulator  64  is similar to that of the modulator  44 . The multiplication  63 ,  73  of the input  61  by the output  65  of an NCO  67  and the output  75  of phase shifter  77  results in a double frequency component and a baseband component in each channel  69 ,  79 . The double frequency component is filtered out by low pass filters (LPF)  83 ,  93  and the outputs  81 ,  91  are just the original baseband signal, if δ, the phase offset in the NCO  67  of the demodulator  64  is small. 
     To simplify the notation we define 
     
       
         ω t+θ=Θ   (1) 
       
     
     Then the result of the top multiplication in FIG. 2 is                             c   1          (   t   )       =                      a   1          (   t   )            cos        (   Θ   )          cos                   (     Θ   +   δ     )       -         a   Q          (   t   )            sin        (   Θ   )            cos        (     Θ   +   δ     )                       =                    1   /   2            a   1          (   t   )            cos        (       2      Θ     +   δ     )         +       1   /   2            a   1          (   t   )          cos                   t        (   δ   )         -                                  1   /   2            a   Q          (   t   )            sin        (       2      Θ     +   δ     )         +       1   /   2            a   Q          (   t   )            sin        (   δ   )                         (   2   )                                  
     and after elimination of the double frequency terms we have 
     
       
           d   I ( t )=½ a   I ( t )cos(δ)+½ a   Q ( t )sin(δ)  (3) 
       
     
     so that for δ=0 we have d I (t)=½ a I (t). 
     Similarly,                             c   Q          (   t   )       =                    -       a   1          (   t   )              cos        (   Θ   )          sin                   (     Θ   +   δ     )       +         a   Q          (   t   )            sin        (   Θ   )            sin        (     Θ   +   δ     )                       =                      -   1     /   2            a   1          (   t   )            sin        (       2      Θ     +   δ     )         -       1   /   2            a   1          (   t   )            sin        (   δ   )         -                                  1   /   2            a   Q          (   t   )            cos        (       2      Θ     +   δ     )         +       1   /   2            a   Q          (   t   )            cos        (   δ   )                         (   4   )                                  
     and after elimination of the double frequency terms, 
     
       
           d   Q ( t )=−½ a   I ( t )sin(δ)+½ a   Q ( t )cos(δ)  (5) 
       
     
     so that for δ=0, d Q (t)=½ a Q (t). 
     Receiver for a Multiple Access Channel 
     In the case of a Spread ALOHA broadcast channel, the results of (3) and (5) show that a synchronous detector for the reception of a signal from a single transmitter can be used for reception. In the case of the multiple access channel, however, the received signal is not in the form of c(t) as defined in FIG. 1, but rather it consists of a sum of signals each having the form of c(t), but each with a different carrier phase and a different modulated signal. Define this received signal as C(t).                C        (   t   )       =         ∑   1   n                         a   kl          (   t   )            cos        (         ω   0        t     +     θ   k       )           -       ∑   1   n                         a   kQ          (   t   )            sin        (         ω   0        t     +     θ   k       )                     (   6   )                                
     where the index k specifies a given transmitter and there are n active transmitters in the interval of interest. 
     Then (3) and (5), which provide the received signal after the high order frequency terms have been filtered out, may be rewritten for the case of the multiple access channel with independent transmitters as,                  D   l          (   t   )       =         1   2            ∑   1   n                         a   kl          (   t   )            cos        (     δ   k     )             +       1   2            ∑   1   n                         a   kQ          (   t   )            sin        (     δ   k     )                       (   7   )                                
     and                  D   Q          (   t   )       =         -     1   2              ∑   1   n                         a   kl          (   t   )            sin        (     δ   k     )             +       1   2            ∑   1   n                         a   kQ          (   t   )            cos        (     δ   k     )                       (   8   )                                
     From (7) and (8), it is shown that in the case of a multiple access channel with asynchronous transmitters (that is, multiple RF phase values), the in-phase and quadrature components of the signal cannot be separated by a single receiver synchronization process. 
     Therefore, the following asynchronous multiple access generalization of the QAM transmitter  40  and receiver  60  described in FIGS. 2 and 3 were considered. We assume that the independent I and Q signals  41 ,  51  of FIG. 1, a I (t) and aQ(t), have been spread by the same spreading sequence, S. 
     The receiver  60  of FIG. 2 is expanded to show the despreading operation at the output of the LPF. In the case of an asynchronous multiple access channel, an identical despreading filter (DS)  85 ,  95  is provided at the output of each channel  81 ,  91  to produce signals  89  and  99 . 
     In order to simplify the analysis, it is assumed that there are only two different active transmitters (k=2). After the problem is solved for the case of k=2, the case for general values of k is obvious. When k=2, equations (7) and (8) become, 
     
       
           D   I ( t )=½ [a   1I ( t )cos(δ 1 )+ a   2I ( t )cos(δ 2 )+ a   1Q ( t )sin(δ 1 )+ a   2Q ( t )sin(δ 2 )]  (9) 
       
     
     and 
     
       
           D   Q ( t )=½ [−a   1I ( t )sin(δ 1 )− a   2I ( t )sin(δ 2 )+ a   1Q ( t )cos(δ 1 )+ a   2Q ( t )cos(δ 2 )]  (10) 
       
     
     Defining the outputs of the two despreading filters as shown in FIG. 3, and then ignoring the scale factor of ½, the equations become, 
     
       
           D   ID ( t )= b   1I ( t )cos(δ 1 )+ b   2I ( t )cos(δ 2 )+ b   1Q ( t )sin(δ 1 )+ b   2Q ( t )sin(δ 2 )+N I   (11) 
       
     
       D   QD ( t )=− b   1I ( t )sin(δ 1 )− b   2I ( t )sin(δ 2 )+ b   1Q ( t )cos(δ 1 )+ b   2Q ( t )cos(δ 2 )+N Q   (12) 
     Equations (11) and (12) provide sequences of samples with both the I samples and the Q samples in both D ID  and D QD . Separation of the desired b**(t) signals by means of an RF phase tracking loop as would be done in the case of a channel with only one received signal is not possible here. Therefore, a suboptimal procedure is provided for detecting the packet, composed of a packet detection procedure followed by a bit detection procedure. 
     Packet Detection/Bit Detection 
     The two channel outputs, D ID  and D QD    89 ,  99  are shown in FIG.  3  and their expressions are provided in equations (11) and (12). This section describes a suboptimal procedure that allows recovery of the bits in the packets received by the asynchronous multiple access channel. Optimality is not claimed for this procedure, only that it will work, it is logical, and the processing required is not onerous. 
     The procedure begins with the process of packet detection, and uses the parameters derived from the packet detection process to make decisions on the bits in the packet. The reason for breaking the process apart in this manner is that the packet detection process operates with an output signal to noise ratio considerably higher than that of the bit detection process. 
     Equations (11) and (12) may be rewritten focusing on a single received signal and, in order to simplify the discussion, introduce a simplified notation. 
     
       
           D   ID ( t )= b   1I ( t )cos(δ 1 )+ b   1Q ( t )sin(δ 1 )+ n   I ( t )  (13) 
       
     
     
       
           D   QD ( t )=− b   1I ( t )sin(δ 1 )+ b   1Q ( t )cos(δ 1 )+ n   Q ( t )  (14) 
       
     
     In order to detect a packet each of the D**(t) signals are passed through two filters matched to the I and Q channel synchronization pulse sequences. The continuous version of these filters have impulse response functions h I (−t) and h Q (−t), respectively. The result of the synching filters is defined on each of the component input signals, as shown in FIG.  4 . 
     The synchronization filters of FIG. 4 are added to the multiple access demodulator, as shown in FIG. 3, to provide the system defined by the block diagram of FIG.  5 . 
     The outputs  89 ,  99  are provided to synchronization filters  103 ,  105 ,  113 ,  115  to produce outputs  101 ,  109 ,  111  and  119 . 
     Then, corresponding to equations (13) and (14), the following equations are obtained, 
     
       
           E   II ( t )= c   1I ( t )cos(δ 1 )+ n   II ( t )  (15) 
       
     
     
       
           E   IQ ( t )= c   1Q ( t )sin(δ 1 )+ n   IQ ( t )  (16) 
       
     
     
       
           E   QI ( t )=− c   1I ( t )sin(δ 1 )+ n   QI ( t )  (17) 
       
     
     
       
           E   QQ ( t )= c   1Q ( t )cos(δ 1 )+ n   QQ ( t )  (18) 
       
     
     Since the packet synchronization sequence is known, the pairs of these signals may then be added  121 ,  123  and then squared  125 ,  127  to obtain an output  131 . 
       E   2   =[E   II ( t )+ E   QQ ( t )] 2   +[E   IQ ( t )− E   QI ( t )] 2   =[c   1I ( t )+ c   1Q ( t )] 2   +N   (19) 
     and packet detection may be accomplished with a threshold on E 2 . Note that the pairs of the signals are added before squaring. This results in a 3 db improvement over simply squaring. 
     Once the start time, T S , of a packet has been determined by the threshold  133  on E 2  it is possible to use that information together with equations (15) through (18) in several ways to estimate either δ 1  or sin(δ 1 ) or cos(δ 1 ). 
     The estimation procedure for sin(δ 1 ) and cos(δ 1 ) is given below. 
     The peak value of the c**(t) occurs at the start time of a packet and is given by, 
     
       
           c **( T   S )=hr  (20) 
       
     
     where h is the length of the packet synch sequence and r is the spreading gain. The estimates of sin(δ 1 ) and cos(δ 1 ) are defined as,                  Est        [     sin        (     δ   1     )       ]       =       S   ^     =     K                       E   IQ     -     E   QI       hr                              (   21   )                                
     and                  Est        [     cos        (     δ   1     )       ]       =       C   ^     =     K                       E   II     +     E   QQ       hr                              (   22   )                                
     where K is a normalization constant to ensure that the sum off the squares of the two estimates given above is equal to one, so that              K   =     hr   E             (   23   )                 S   ^     =                    E   IQ     -     E   QI       E             (   24   )                                
     and                  C   ^     =                    E   II     -     E   QQ       E                          (   25   )                                
     Once the start time of a packet has been determined the bit values of the bits in the packet can be obtained by using (24) and (25), as if they were known values, together with equations (13) to (14). Equations (13) and (14) can be rewritten as 
     
       
           D   ID ( t )= Ĉb   1I ( t )+ Ŝb   1Q ( t )+ n   I ( t )  (26) 
       
     
     
       
           D   QD ( t )=− Ŝb   1I ( t )+ Ĉb   1Q ( t )+ n   Q ( t )  (27) 
       
     
     Solving these equations for the b**(t) we get 
     
       
           b   1I ( t )= ĈD   ID ( t )− ŜD   QD ( t )+ n   1 ( t )  (28) 
       
     
     
       
           b   IQ ( t )= ŜD   ID ( t )+ ĈD   QD ( t )+ n   2 ( t )  (29) 
       
     
     If the n I (t) and n Q (t) terms are independent zero mean Gaussian random variables with variances equal to V 2  then so are the n 1 (t) and n 2 (t) terms. Then, using (28), the minimum probability of error decision rule for determining whether b 1I  is positive or negative is given by 
     
       
           {circumflex over (b)}   1I   =r  if  ĈD   ID ( t )− ŜD   QD &gt;0 
       
     
     
       
           {circumflex over (b)}   1I   =−r  if  ĈD   ID ( t )− ŜD   QD &lt;0  (30) 
       
     
     A similar argument starting from (29) leads to a similar binary hypothesis test for the Q channel 
     
       
           {circumflex over (b)}   1Q   =r  if  ŜD   ID ( t )+ ĈD   QD ( t )&gt;0 
       
     
     
       
           {circumflex over (b)}   1Q   =−r  if  ŜD   ID ( t )+ ĈD   QD ( t )&lt;0  (31) 
       
     
     Equations (30) and (31) are not necessarily optimum decisions because the estimated values of the sine and cosine were used as if they were true values, and because it was assumed that the noise in (26) and (27) is Gaussian. But both assumptions look good because the signal to noise ratio out of the packet detection filter is much better than that out of the bit detection filter. 
     When the expressions for S and C are used in the inequalities (30) and (31) the following equations are obtained, 
     
       
           {circumflex over (b)}   1I   =r  if  D   ID ( E   II   +E   QQ )− D   QD ( E   IQ   −E   QI )&gt;0 
       
     
     
       
           {circumflex over (b)}   1I   =−r  if  D   ID ( E   II   +E   QQ )− D   QD ( E   IQ   −E   QI )&lt;0  (32) 
       
     
     and 
     
       
           {circumflex over (b)}   1Q   =r  if  D   ID ( E   IQ   −E   QI )+ D   QD ( E   II   +E   QQ )&gt;0 
       
     
     
       
           {circumflex over (b)}   1Q   =−r  if  D   ID ( E   IQ   −E   QI )+ D   QD ( E   II   +E   QQ )&lt;0  (33) 
       
     
     While the invention has been described with reference to specific embodiments, modifications and variations of the invention may be constructed without departing from the scope of the invention, which is defined in the following claims.