Abstract:
A frequency comparator includes a circuit comparing, independently of a phase relationship between first and second clocks, frequencies of the first and second clocks and outputting first and second detection signals when the first clock has frequencies higher and lower than those of the second clock, respectively. The first and second detection signals are output for respective times based on a difference between the frequencies of the first and second clocks.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    The present invention generally relates to frequency comparators and clock regenerating devices using the same, and more particularly to a frequency comparator which compares digital signals and a clock regenerating device using such a frequency comparator.  
           [0003]    2. Description of the Related Art  
           [0004]    A clock reproduction device is known which uses a PLL (Phase-Locked Loop) and regenerates a clock from input data. In such a clock regenerating device, an output clock of the PLL is compared with a reference clock, and a voltage-controlled oscillator (VCO) provided in the PLL is controlled based on an error which corresponds to the difference between the output clock and the reference clock.  
           [0005]    A clock regenerating device as described above is disclosed in U.S. Pat. No. 5,015,970. The clock regenerating device frequency-divides the output signal of the VCO at frequency-dividing ratios of 1/(N+1) and 1/(N−1). The two signals thus obtained are phase-compared with a reference clock obtained by frequency-dividing the output clock of the VCO at a frequency-dividing ratio of 1/M, so that a coarse adjustment signal can be obtained. The output clock of the VCO is also phase-compared with input data, and thus a fine adjustment is obtained. The coarse adjustment signal and the fine adjustment signals are added by a smoothing filter, and a resultant control signal which controls the VCO is obtained.  
           [0006]    However, the above clock regenerating device has the following disadvantages. First, the device obtains the coarse adjustment signal by the phase comparing operation, and does not perform a frequency comparing operation independent of phases. Second, the device generates the VCO control signal which changes the phase of the output clock of the VCO on the basis of the coarse adjustment signal, and thus does not operate stably so that the PLL attempts to obtain a plurality of different in-phase (phase synchronized) states. If it is attempted to avoid unstable operation, there is nothing other than a modification in which the coarse adjustment is carried out more slowly. However, it takes a longer time to complete the coarse adjustment.  
         SUMMARY OF THE INVENTION  
         [0007]    It is a general object of the present invention to provide a frequency comparator and a clock regenerating device using the same in which the above disadvantages are eliminated.  
           [0008]    A more specific object of the present invention is to provide a frequency comparator capable of comparing frequencies independent of phases and generating a coarse adjustment signal by frequency comparison so that a stable and high-speed frequency pull-in operation can be carried out and to provide a clock reproduction device using such a frequency comparator.  
           [0009]    The above objects of the present invention are achieved by a frequency comparator comprising: a circuit comparing, independently of a phase relationship between first and second clocks, frequencies of the first and second clocks and outputting first and second detection signals when the first clock has frequencies higher and lower than those of the second clock, respectively, the first and second detection signals being output for respective times based on a difference between the frequencies of the first and second clocks. Hence, it is possible to detect the frequency difference independently of the phase relationship between the first and second clocks and to thus obtain the first and second detection signals having pulse widths corresponding to the frequency difference.  
           [0010]    The frequency comparator may be configured so that the circuit outputs the first detection signal when the frequency of the first clock is higher than that of the second clock at a first ratio or more, and outputs the second detection signal when the frequency of the first clock is lower than that of the second clock at a second ratio or more. Hence, it is possible to define an insensitive range in which none of the first and second detection signals are output when the first and second clocks have respective frequencies close to each other.  
           [0011]    The frequency comparator may be configured so that the circuit comprises: a first counter which counts the first clock and outputs the first detection signal when a first count value of the first counter is equal to or greater than (n+a) where n and a are integers; a second counter which counts the second clock and outputs the second detection signal when a second count value of the second counter is equal to or greater than (n+b) where b is an integer; and a reset unit which resets the first and second counters when the first and second count values are both equal to or greater than the n. Hence, it is possible to detect the frequency difference independently of the phase relationship between the first and second clocks and to thus obtain the first and second detection signals having pulse widths corresponding to the frequency difference.  
           [0012]    The frequency comparator may be configured so that one of the first and second detection signals is output until the reset unit resets the first and second counters.  
           [0013]    The frequency comparator may be configured so that the circuit comprises: a first counter which counts the first clock and outputs the first detection signal when a first count value of the first counter is equal to or greater than (n+a) where n and a are integers; a second counter which counts the second clock and outputs the second detection signal when a second count value of the second counter is equal to or greater than (n+b) where b is an integer; a reset unit which generates a reset signal which resets the first and second counters when the first and second count values are both equal to or greater than the n; a first hold circuit which holds the first detection signal until the first hold circuit is reset in response to the reset signal; and a second hold circuit which holds the second detection signal until the second hold circuit is reset in response to the reset signal. Hence, it is sufficient that the first and second counters can count up to (n+a) and (n+b), respectively, so that the counters can be simplified.  
           [0014]    The above objects of the present invention are also achieved by a clock regenerating device comprising: a frequency comparator comparing, independently of a phase relationship between first and second clocks, frequencies of the first and second clocks and outputting first and second detection signals when the first clock has frequencies higher and lower than those of the second clock, respectively, the first and second detection signals being output for respective times based on a difference between the frequencies of the first and second clocks; a phase comparator generating a fine-adjustment signal based on a difference between a phase of input data and a phase of the second clock; a combining unit generating a control signal from the first and second detection signals serving as a coarse-adjustment signal and the fine-adjustment signal; and an oscillator outputting a regenerated clock which is an oscillation output having a frequency based on the control signal and corresponds to the second clock. The combining unit combining the first and second signals and the fine-adjustment signal so that the following condition is satisfied: 
           (Δ pf/Δff )&lt;(Δ pc/Δfc ) 
           [0015]    where Δpc and Δfc respectively denote a phase variation and a frequency variation in the regenerated clock caused by the coarse-adjustment signal per unit time, and Δpf and Δff respectively denote a phase variation and a frequency variation in the regenerated clock caused by the fine-adjustment signal per unit time. Hence, it is possible to independently obtain frequency information from the second clock and phase information from the input data without any interference and to rapidly pull the circuit in phase.  
           [0016]    The clock regenerating device may be configured so that the fine-adjustment signal includes fine-adjustment up and down signals based on the phase relationship between the input data and the second clock.  
           [0017]    The clock regenerating device may be configured so that the coarse-adjustment signal is output during only a time based on the frequency difference between the first clock and the second clock.  
           [0018]    The clock regenerating device may be configured so that the coarse-adjustment signal includes a voltage or current based on the frequency difference between the first clock and the second clock.  
           [0019]    The clock regenerating device may be configured so that the frequency comparator comprises: a first counter which counts the first clock and outputs the first detection signal when a first count value of the first counter is equal to or greater than (n+a) where n and a are integers; a second counter which counts the second clock and outputs the second detection signal when a second count value of the second counter is equal to or greater than (n+b) where b is an integer; and a reset unit which resets the first and second counters when the first and second count values are both equal to or greater than the n.  
           [0020]    The clock regenerating device may be configured so that the combining unit comprises an adder unit which adds an integrated value of the coarse-adjustment signal, an integrated value of the fine-adjustment signal, and an instantaneous value of the fine-adjustment signal. Hence, it is possible to integrate the coarse-adjustment and fine-adjustment signals by means of a common integrator and to completely eliminate the phase information from the second clock originally unnecessary in the in-phase pulling operation.  
           [0021]    The clock regenerating device may be configured so that the combining unit comprises an adder unit which adds the coarse-adjustment signal and the fine-adjustment signal; an integral unit which integrates an output signal of the adder unit; an instantaneous voltage generating unit which generates an instantaneous voltage based on the fine-adjustment signal; and a mixing unit which adds an output of the integral unit and the instantaneous voltage and thus generates the control signal. Hence, it is possible to obtain the control signal based on both the frequency comparing operation and the phase comparing operation.  
           [0022]    The clock regenerating device may be configured so that the adder unit comprises: a first current switch which charges a first capacitor provided in the integral unit with a constant current by a supply of the first detection signal; a second current switch which discharges the first capacitor with a constant current by a supply of the second detection signal; a third current switch which charges the first capacitor with a constant current by a supply of the first-adjustment signal; and a fourth current switch which discharges the first capacitor with a constant current by a supply of the fine-adjustment signal. Hence, it is possible to perform the integral operation by charging and discharging the first capacitor on the basis of the first and second detection signals and the fine-adjustment signal.  
           [0023]    The clock regenerating device may be configured so that the adder unit comprises: a first OR circuit which performs an OR operation on the first detection signal and the fine-adjustment signal and thus outputs a first output; a second OR circuit which performs an OR operation on the second detection signal and the fine-adjustment signal and thus outputs a second output; a fifth switch which charges a first capacitor provided in the integral unit with a constant current by the first output; and a sixth switch which discharges the first capacitor with a constant current by the second output. Hence, it is possible to perform the integral operation by charging and discharging the first capacitor on the basis of the first and second detection signals and the fine adjustment signal.  
           [0024]    The clock regenerating device may be configured so that the mixing unit comprises: a buffer supplied with the output of the integral unit; and a first resistor supplied with an output of the buffer, the instantaneous voltage generating unit supplying a constant current to the first resistor in response to a supply of the fine-adjustment signal. Hence, the control signal can be obtained by adding the voltage generated by the constant current flowing in the first resistor in accordance with a supply of the fine-adjustment signal.  
           [0025]    The clock regenerating device may be configured so that the instantaneous voltage generating unit comprises: a second resistor; a constant-voltage source which applies a constant voltage across the second resistor; and a unit which causes a current proportional to a current flowing in the second resistor to flow in the first resistor by a supply of the fine-adjustment signal. Hence, it is possible to stabilize the voltage drop developed across the first resistor by the current based on a supply of the fine-adjustment signal independent of variations in temperature, power supply and/or production process.  
           [0026]    The clock regenerating device may be configured so that: the instantaneous voltage generating unit comprises an inverted amplifier having the first resistor as a feedback element; and the integral unit comprises another inverted amplifier having the first capacitor as a feedback element.  
           [0027]    The clock regenerating device may be configured so that it further comprises a voltage dividing circuit which includes resistors connected in series and has an end fixed to a given potential, a divided voltage being supplied to the oscillator. Hence, it is possible to reduce the capacitance of the first capacitor and reduce the chip area occupied by the clock regenerating circuit.  
           [0028]    The clock regenerating device may further comprise a phase-locked loop which is formed on a semiconductor chip on which the clock regenerating device is formed and which includes another integral unit which outputs said given potential. The two integral units are formed on the same chip and the first capacitor can further be reduced.  
           [0029]    The clock regenerating device may be configured so that the oscillator comprises: a charge current switch supplying a charging current; a discharge current switch supplying a discharge current; a second capacitor which is charged and discharged by the charging and discharging currents; and a circuit part which generates a switching signal for turning ON one of the charge current switch and the discharge current switch by comparing a voltage developed across the second capacitor with first and second reference voltages and generating the regenerated clock from the switching signal, the charge current switch and the discharge current switch being coupled to the second capacitor through respective current output transistors. Hence, it is possible to suppress occurrence of noise caused by switching of the current charge and discharge switches. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0030]    Other objects, features and advantages of the present invention will become more apparent from the following detailed description when read in conjunction with the accompanying drawings, in which:  
         [0031]    [0031]FIG. 1 is a block diagram of a frequency comparator according to an embodiment of the present invention;  
         [0032]    [0032]FIG. 2 is a timing chart of an operation of the frequency comparator shown in FIG. 1;  
         [0033]    [0033]FIG. 3 is a timing chart of another operation of the frequency comparator shown in FIG. 1;  
         [0034]    [0034]FIG. 4 is a timing chart of yet another operation of the frequency comparator shown in FIG. 1;  
         [0035]    [0035]FIG. 5 is a circuit diagram of the frequency comparator shown in FIG. 1;  
         [0036]    [0036]FIG. 6 is a block diagram of a clock regenerating device according to the present invention;  
         [0037]    [0037]FIG. 7 is a timing chart of an operation of the clock regenerating device shown in FIG. 6;  
         [0038]    [0038]FIG. 8 is a block diagram of the clock regenerating device shown in FIG. 6 in more detail;  
         [0039]    [0039]FIG. 9 is a circuit diagram of a phase comparator;  
         [0040]    [0040]FIG. 10 is a circuit diagram of a filter unit;  
         [0041]    [0041]FIG. 11 is a circuit diagram of current switch;  
         [0042]    [0042]FIG. 12 is a circuit diagram of another configuration of the filter unit;  
         [0043]    [0043]FIG. 13 is a block diagram of another clock regenerating device;  
         [0044]    [0044]FIG. 14A is an equivalent circuit diagram of an adder unit, an integral unit and a mixing unit shown in FIG. 13;  
         [0045]    [0045]FIG. 14B is an equivalent circuit diagram of an integral unit and a mixing unit shown in FIG. 14A;  
         [0046]    [0046]FIG. 15 is a waveform diagram showing the principle of reduction in a capacitor shown in FIG. 14A;  
         [0047]    [0047]FIG. 16 is a circuit diagram of a voltage controlled oscillator;  
         [0048]    [0048]FIG. 17 is a timing chart of an operation of the filter unit shown in FIG. 12;  
         [0049]    [0049]FIG. 18 is a circuit diagram of a current switch;  
         [0050]    [0050]FIG. 19 is a circuit diagram of another configuration of the current switch; and  
         [0051]    [0051]FIGS. 20A and 20B show the principle of a charge pump type low-pass filter. 
     
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0052]    [0052]FIG. 1 is a block diagram of a frequency comparator according to an embodiment of the present invention. A reference clock (frequency f 0 ) serving as a first clock is applied to a first counter  12  via a terminal  10 . A compared clock (frequency f 1 ) serving as a second clock is applied to a second counter  16  via a terminal  14 . The first counter  12  is reset to zero in response to a reset signal, and then starts counting the reference clock. When the count value becomes equal to n, the first counter  12  supplies an n detection signal to a reset circuit  18 . When the count value becomes equal to n+a, the first counter  12  supplies an (n+a) detection signal to a first hold circuit  20 . The second counter  16  is reset to zero in response to the reset signal, and then starts counting the supplied clock. When the count value of the second counter  16  becomes equal to n, the second counter  16  supplies an n detection signal to the reset circuit  18 . When the count value becomes equal to n+b, the second counter  16  supplies an (n+b) detection signal to a second hold circuit  22 .  
         [0053]    The reset circuit  18  generates the reset signal when receiving the n detection signals from both the first counter  12  and the second counter  16 , so that the first and second counters  12  and  16  are reset to zero and the first and second hold circuits  20  and  22  are reset. When the first hold circuit  20  is supplied with the (n+a) detection signal from the first counter  12 , the first hold circuit  20  outputs, to a terminal  24 , a first detection signal, which is, for example, at a high level, until it is reset. When the second hold circuit  22  is supplied with the (n+b) detection signal from the second counter  16 , the second hold circuit  22  outputs, to a terminal  26 , a second detection circuit, which is, for example, at a high level, until it is reset.  
         [0054]    In a case where a condition, n·f 0 /(n+a)≦f 1 ≦(n+b) is satisfied, one of the first and second counters  12  and  16  supplied with the reference clock and the compared clock respectively shown in FIGS.  2 (A) and  2 (B) detects the count value n ahead of the other. Then, the other counter detects the count value n before (n+a) or (n+b) is detected. The reset circuit  18  performs the reset operation when both the counters  12  and  16  detect the count value n, and thus the first and second detection signals are not output, as shown in FIGS.  2 (C) and  2 (D), respectively.  
         [0055]    In a case where a condition, f 1 &gt;(n+b)·f 0 /n is satisfied, the second counter  16  detects the count value n by the compared clock shown in FIG. 3(B), and then detects the count value (n+b). In this case, b is equal to 1. Hence, a shown in FIG. 3(D), the second detection signal which is at the high level is generated. Thereafter, the first counter  12  detects the count value n by the reference clock shown in FIG. 3(A). Thus, the reset circuit  18  performs the reset operation, and the second detection signal is switched to the low level, as shown in FIG. 3(D). In this case, the first detection signal is not output, as shown in FIG. 3(C).  
         [0056]    In short, when the frequency f 1  of the clock is higher than that of the reference clock f 0  by a given value or more, the second detection signal is output. The output period (pulse width) of the second detection signal can be written as n/f 0 −(n+b)/f 1 .  
         [0057]    In a case where a condition, f 1 &lt;n·f 0 /(n+a) is satisfied, the first counter  12  detects the count value n by the reference clock, and then detects the count value (n+a). In this case, a is equal to 1. Hence, as shown in FIG. 4(C), the first detection signal that is at the high level is generated. Then, when the second counter  16  detects the count value n by the compared clock shown in FIG. 4(B), the reset circuit  18  performs the reset operation, and the first detection signal is switched to the low level, as shown in FIG. 4(C). In this case, the second detection signal is not output, as shown in FIG. 4(D).  
         [0058]    In short, when the frequency f 1  of the clock is lower than that of the reference clock f 0  by a given value or more, the first detection signal is output. The output period (pulse width) of the first detection signal can be written as n/f 1 −(n+a)/f 0 .  
         [0059]    [0059]FIG. 5 is a circuit diagram of the frequency comparator shown in FIG. 1. In FIG. 5, parts that are the same as those shown in FIG. 1 are given the same reference numbers. The first counter  12  is supplied with the reference clock, and includes counters  30  and  31 , AND gate  32  and inverters. A carry signal of the counter  30  which is output via its carry-out terminal is applied to a carry-in terminal of the counter  31 . When n is equal to  126 , the n detection signal which is at the high level is output by the AND circuit  32 . When (n+a) is equal to  128 , the (n+a) detection signal which is at the high level is output via a data output terminal Q of the counter  31 . The second counter  16  is supplied with the compared clock, and is made up of counters  35  and  36 , an AND circuit  37  and inverters. A carry signal of the counter  35  which is output via its carry-out terminal is applied to a carry-in terminal of the counter  36 . When n is equal to 126, the n detection signal which is at the high level is output by the AND circuit  37 . When (n+a) is equal to 128, the (n+a) detection signal which is at the high level is output via a data output terminal Q D  of the counter  36 .  
         [0060]    The reset circuit  18  is made up of D-type flip-flops  40  and  41 , a NAND circuit  42 , and a delay circuit  43 . The flip-flops  40  and  41  respectively latch the n detection signals from the AND circuits  32  and  37  in synchronism with signals obtained by inverting the reference clock and the compared clock by inverters  33  and  38 . This is intended to avoid hazard. The NAND circuit  42  generates the reset signal which is at the low level when the signals of the flip-flops  40  and  41  are both switched to the high level. The reset signal thus generated resets the counters  30 ,  31 ,  35  and  37  to zero, and resets D-type flip-flops  46  and  48  of the first and second hold circuits  20  and  22 . The reset signal is delayed by a given time by the delay circuit  43 , and the delayed signal resets the flip-flops  40  and  41  after the counters  30 ,  31 ,  35  and  37  are reset.  
         [0061]    The first hold circuit  20  includes D-type flip-flops  45  and  46 . The flip-flop  45  latches the (n+a) detection signal in synchronism with the output clock of the inverter  33 . The flip-flop  46  receives the output signal of the flip-flop  45  as a clock input, and continuously receives a high-level data signal. The output signal of the flip-flop  46  is output via the terminal  24  as the first detection signal.  
         [0062]    The second hold circuit  22  includes D-type flip-flops  47  and  48 . The flip-flop  47  latches the (n+b) detection signal in synchronism with the output clock of the inverter  38 . The flip-flop  48  receives the output signal of the flip-flop  47  as a clock input, and continuously receives a high-level data signal. The output signal of the flip-flop  48  is output via the terminal  25  as the second detection signal.  
         [0063]    In the above-mentioned embodiment of the present invention, when the difference between the frequency f 0  and the frequency f 1  exceeds the given value (equal to a or b), the first and second detection signals having the pulse widths corresponding to the frequency difference are obtained, and are not affected by the phase difference between the signals of the frequencies f 0  and f 1 .  
         [0064]    If the first and second counters  12  and  16  are saturated at the respective count values (n+a) and (n+b), the first and second hold circuits  20  and  22  will be unnecessary. For example, the counters  12  and  16  are formed of 8-bit binary counters. If (n+a)=(n+b)=128, the 2″ outputs of the binary counters are equal to 1 when the count values are equal to any of 128-255. Hence, as long as f 0 /2&lt;f 1 &lt;2f 0  is satisfied, the binary counters can be considered as being of a saturation type.  
         [0065]    In the above-mentioned embodiment of the present invention, the given values a and b are equal to 2 so that an insensitive range can be defined when the count values of the first and second counters  12  and  16  fall within the range from (n+a−1) to (n+b−1). As long as the count values fall within the insensitive range, the first and second detection signals are not output. Alternatively, it is possible to set the given values a and b equal to 0 and to thus employ no insensitive range.  
         [0066]    [0066]FIG. 6 is a block diagram of a frequency comparator according to another embodiment of the present invention. A reference clock serving as a first clock shown in FIG. 7(A) is applied to a terminal  200 , and is then applied to an n-ary counter  202 . A compared clock serving as a second clock shown in FIG. 7(B) is applied to a terminal  204 , and is then applied to a counter  206 . The counter  202  counts the reference clock and outputs a carry each time the count value becomes equal to n. The above carry is supplied, as a reset signal, to the counter  206 . The counter  206  counts the compared clock, and supplies a D/A converter  208  with a digital value which is equal to the difference (n−m) where m denotes the count value obtained when the reset signal is supplied thereto. The D/A converter  208  converts the digital value (n−m) into an analog signal at the time of supply of the reference clock as shown in FIG. 7(C). The analog signal thus obtained is output via a terminal  210  as the first detection signal. The second detection signal can be generated in the same manner as described above.  
         [0067]    [0067]FIG. 8 is a block diagram of a clock regenerating device equipped with the frequency comparator shown in FIG. 1. A frequency comparator (FD)  50  shown in FIG. 8 has the configuration shown in FIG. 1. The reference clock applied to a terminal  52  is supplied to the first counter  12 . The second counter  16  is supplied with the clock generated by a voltage-controlled oscillator (VCO)  54 , which will be described later. The first and second detection signals respectively output by the first and second hold circuits  20  and  22  are supplied, as a coarse-adjustment up signal and a coarse-adjustment down signal, to an adder unit  62  provided in a combining circuit  60 .  
         [0068]    A phase comparator (PD)  56  compares the phase of the output clock from the VCO  54  with the phase of input data incoming via a terminal  58 , and a fine-adjustment up signal and a fine-adjustment down signal to the adder unit  62  and an instantaneous voltage generating unit  64  provided in the combining unit  60 . The adder unit  62  subtracts the coarse-adjustment down signal and the fine-adjustment down signal from the coarse-adjustment up signal and the fine-adjustment up signal, and supplies the resultant signals to an integral unit  66 . The integral unit  66  integrates the two output signal voltages from the adder unit  62 . The integrated voltage corresponds to a frequency control component for the VCO  54 , and is supplied to a mixing unit  67 . The instantaneous voltage generating unit  64  generates an instantaneous voltage as the frequency control component for the VCO  54 , the instantaneous voltage functioning to increase the voltage in accordance with the fine-adjustment up signal and decrease the voltage in accordance with the fine-adjustment down signal. The above instantaneous voltage is supplied to the mixing unit  67 . The mixing unit  67  mixes (adds) the integrated voltage output by the integral unit  66  and the instantaneous voltage output by the instantaneous voltage generating unit  64 , and supplies a resultant control voltage to the VCO  54 . The VCO increases the oscillation frequency as the control voltage increases so as to generate the clock having the frequency and phase corresponding to the control voltage. The clock thus generated is output via a terminal  68 . The coarse-adjustment up and down signals have a phase response weaker than that of the fine-adjustment up and down signals, and have a frequency response stronger than that of the fine-adjustment up and down signals. The above means that the following is satisfied: 
         Δ pf/Δff&lt;Δpc/Δfc   
         [0069]    where Δpc and Δfc respectively denote a phase variation and a frequency variation in the output of the VCO  54  due to activation (the high level in the embodiment being concerned) of the coarse-adjustment up and down signals per unit time, and Δpf and Δff respectively denote a phase variation and a frequency variation in the output of the VCO  54  due to activation (the high level in the embodiment being concerned) of the fine-adjustment up and down signals per unit time.  
         [0070]    The phase variation results from a frequency variation which takes place while the coarse-adjustment or fine-adjustment up and down signals are activated. The frequency variation results from a variation in the output of the integral unit  66  which takes places while the coarse-adjustment or fine-adjustment up and down signals are activated.s, and is maintained even after the above signals are inactivated.  
         [0071]    [0071]FIG. 9 is a circuit diagram of the phase comparator  56 . The input data is applied to AND circuits  73  and  74  via a terminal  71 . The output clock of the VCO  54  is applied to a terminal  72 , and is supplied to the AND circuit  74  and an inverter  75 . The inverted signal from the inverter  75  is supplied to the AND circuit  73 .  
         [0072]    If the phase of the input data leads to that of the clock, the AND circuit  73  generates a pulse having a pulse width corresponding to the phase difference from the rising edge of the input data to the rising edge of the clock. The pulse thus generated is output via a terminal  76  as the fine-adjustment up signal. The AND circuit  74  generates a pulse having a pulse width corresponding to the phase difference from the rising edge of the clock to the falling edge of the input data. The pulse thus generated is output via a terminal  77  as the fine-adjustment down signal.  
         [0073]    [0073]FIG. 10 is a circuit diagram of the combining unit  60 . The coarse-adjustment up signal and the coarse-adjustment down signal are respectively applied to terminals  81  and  82 , and are supplied to first and second current switches  83  and  94  provided in the adder unit  62 . The fine-adjustment up signal and the fine-adjustment down signal are respectively applied to terminals  85  and  86 , and are supplied to third and fourth current switches  87  and  88  provided in the adder unit  62  and fifth and sixth current switches  89  and  90  provided in the instantaneous voltage generating unit  64 .  
         [0074]    The first current switch  83  inverts the coarse-adjustment up signal, and supplies the inverted signal to the gate of a P-channel MOS transistor PT 1 . A capacitor C of the integral unit  66  is charged by a constant-current source ip 1  during the high-level period of the coarse-adjustment up signal. The second current switch  84  supplies the coarse-adjustment down signal to the gate of an N-channel MOS transistor NT 1 . The capacitor C is discharged by a constant-current source in 1  during the high-level period of the coarse-adjustment down signal.  
         [0075]    The third current switch  87  inverts the fine-adjustment up signal, and supplies the inverted signal to the gate of a P-channel MOS transistor PT 2 . The capacitor C is charged by a constant-current source ip 2  during the high-level period of the fine-adjustment up signal. The fourth current switch  88  supplies the fine-adjustment down signal to the gate of an N-channel MOS transistor NT 2 . The capacitor C 1  is discharged by a constant-current source in 2  during the high-level period of the fine-adjustment down signal.  
         [0076]    The fifth current switch  89  inverts the fine-adjustment up signal, and supplies the inverted signal to the gate of a P-channel MOS transistor PT 3 . Hence, a current flows, during the high-level period of the fine-adjustment up signal, in a resistor R of the integral unit  66  by a constant-current source ip 3  so that the output voltage is increased. The sixth current switch  90  supplies the fine-adjustment down signal to the gate of an N-channel MOS transistor NT 4 . A current flows, during the high-level period of the fine-adjustment down signal, in the resistor R by a constant-current source in 3  so that the output voltage is decreased.  
         [0077]    The integral unit  66  includes the capacitor C, which integrates the signal supplied from the adder unit  62 . The capacitor C is connected to a buffer  91  of a source-follower structure made up of the N-channel MOS transistor NT 4  and the constant-current source in 4 . The buffer  91  functions as the mixing unit  67 . Hence, the voltage developed across the capacitor C is not affected when the current switches  89  and  90  of the instantaneous voltage generating unit  64  cause the current to flow in the resistor R. Hence, the VCO control signal which corresponds to the sum of the output voltage of the buffer  91  and the voltage developed across the resistor R in which the current flows can be obtained at a terminal  92 .  
         [0078]    The VCO control voltages now labeled VCNT respectively obtained by the embodiment of the present invention and the prior art can be expressed as follows: 
           V   CNT   =I   F   ·R+ (1/ C )∫( I   C   +I   F ) dt   (1) 
           V   CNT =( I   C   +I   F )· R+ (1/ C )∫( I   C   +I   F ) dt   (2) 
         [0079]    where I C  denotes the current generated by the current switches  83  and  84  by the coarse-adjustment up and down signals, and I F  denotes the current generated by the current switches  87  and  88  or  89  and  90  by the fine-adjustment up and down signals. The VCO control voltage expressed by equation (1) is generated by the embodiment of the present invention. The VCO control voltage expressed by equation (2) is generated by the prior art.  
         [0080]    The first term of equation (1) is an instantaneous value as a phase control component depending on only the fine adjustment signals, and the second term thereof is a frequency control component depending on the integrated value of the sum of the coarse adjustment signals and the fine adjustment signals. The present invention is directed to pulling the regenerated clock output by the VCO  54  in phase with the input data, and is not required to pull the regenerated clock in phase with the reference clock. This is because there is no phase relationship between the input data and the reference clock. It can be seen from equation (1) that the above object of the present invention is achieved.  
         [0081]    Equation (2) indicating the operation of the prior device differs from equation (1) in that the first term of equation (2) depends on the sum of the coarse adjustment signals and the fine adjustment signals. That is, there is no phase relationship between the input data and the reference clock, nevertheless the device operates so that the regenerated clock is pulled in phase with the reference clock. Hence, the above operation prevents the regenerated clock from being pulled in phase with the input data.  
         [0082]    The constant-current sources ip 3  and in 3  of the current switches  89  and  90  may be configured as shown in FIG. 11. In FIG. 11, a stabilized reference voltage is supplied to the non-inverting input terminal of an operational amplifier  95 , and the output signal thereof is applied to the gate of an N-channel MOS transistor  96 . The source of the transistor  96  is grounded via a resistor Rc formed in a semiconductor chip on which the clock regenerating device is formed, and is connected to the inverting input terminal of the operational amplifier  95 . The drain current of the transistor  96  is supplied from a P-channel MOS transistor  97 . The transistor  97  forms current-mirror circuits in cooperation with the constant-current sources ip 3  and in 3 .  
         [0083]    If the resistor R has a resistance deviation, the resistor Rc formed on the same chip as the resistor R will have an identical resistance deviation. The operational amplifier  95  controls the drain current of the transistor  96  so that the voltage drop developed across the resistor Rc is constant. The above control of the drain current varies the magnitudes of the currents of the constant-current sources ip 3  and in 3 . The voltage drops developed across the resistor R when the currents of the constant-current sources ip 3  and in 3  respectively flow through the resistor R is made constant irrespective of variations and deviations in temperature, power supply voltage and/or production process. The first, second, third and fourth current switches  83 ,  84 ,  87  and  88  employ constant-current sources independent of the resistor Rc. With the above-mentioned arrangement, the performance of the PLL loop characteristics can be stabilized and the clock can stably be regenerated.  
         [0084]    [0084]FIG. 12 is a circuit diagram of a variation of the combining unit  60 . The coarse-adjustment up signal and the coarse-adjustment down signal are respectively applied to terminals  81  and  82 , and are supplied to current switches  100  and  101  forming an instantaneous voltage generating circuit, and to OR circuits  104  and  105  forming an adder unit. The fine-adjustment up signal and the fine-adjustment down signal are respectively applied to terminals  85  and  86 , and are supplied to current switches  102  and  103  forming the instantaneous voltage generating circuit  64  and OR circuits  106  and  107 .  
         [0085]    The fifth current switch  106  inverts the coarse-adjustment up signal, and supplies the inverted signal to the gate of the P-channel MOS transistor PT 1 . A constant-current source ip 4  charges the capacitor C of the integral unit  66  during the high-level period of the coarse-adjustment up signal or the fine-adjustment up signal. The sixth current switch  107  supplies the coarse-adjustment down signal to the gate of the N-channel MOS transistor in 4 , and the capacitor C is discharged by the constant-current source in 4  during the high-level period of the coarse-adjustment down signal or the fine-adjustment down signal.  
         [0086]    The current switches  100  and  102  respectively invert the coarse-adjustment and fine-adjustment up signals, and applies the inverted signals to the gates of respective P-channel MOS transistors. Hence, currents flow, during the high-level periods of the signals, through the resistor R of the integral unit  66  by constant-current sources ip 5  and ip 6  so that the output voltage is increased. The current switches  101  and  103  supply the coarse-adjustment and fine-adjustment down signals to the gates of respective N-channel MOS transistors. Hence, currents flow, during the high-level periods of the signals, through the resistor R by constant-current sources in 5  and in 6  so that the output voltage is decreased.  
         [0087]    The currents of the constant-current sources have the following relationships: 
         
       ip 
       4 
       =in 
       4 
     
           ip   5   =in   5   &lt;ip   6   =in   6 . 
         [0088]    In the present variation, the frequency variations commonly own the constant-current sources  106  and  107  for the coarse and fine adjustments, and the following relationships stand: 
         Δ ff=Δfc   
         Δ pf&lt;Δpc.   
         (Δ pf/Δff )&lt;(Δ pc/Δfc ). 
         [0089]    It will be noted that a large area for forming the capacitor C of the integral unit  66  will be occupied on the semiconductor chip. FIG. 13 shows an embodiment of the present invention directed to reducing the capacitor C (the capacitance thereof). In FIG. 13, parts that are the same as those shown in FIG. 10 are given the same reference numbers.  
         [0090]    Referring to FIG. 13, the output terminal of the integral unit  66  is connected to a resistor R 1 , to which a resistor R 2  is connected, so that a voltage dividing circuit is thus formed. The resistor R 2  is connected to an output terminal of an adder/integral unit  111 . A connection node at which the resistors R 1  and R 2  are connected in series is connected to the VCO  54 . A phase/frequency comparator (PFD)  110  compares the phase and frequency of the reference clock supplied via the terminal  52  with those of the clock supplied from a frequency divider  114 , and outputs coarse-adjustment and fine-adjustment up and down signals to the integral unit  111 . The fine-adjustment up and down signals from the comparator  110  are also supplied to an instantaneous voltage generating circuit  112 . The adder/integral unit  111  adds the coarse-adjustment and fine-adjustment up and down signals and also integrates these signals. The instantaneous voltage generating circuit  112  generates an instantaneous voltage in accordance with the fine-adjustment up and down signals. A mixing unit  115  mixes the integrated value output by the adder/integral unit  111  with the instantaneous voltage, and thus generates a VCO control signal. The above-mentioned comparator  110 , the adder/integral unit  111  and the instantaneous voltage generating circuit  112  respectively have the same structures as those of the aforementioned frequency comparator (FD)  50 , the phase comparator (PD)  56 , the instantaneous voltage generating circuit  64 , the adder unit  62  and the integral unit  66 .  
         [0091]    The VCO control signal output by the adder/integral unit  111  is supplied to a VCO  113 , which generates a sampling clock synchronized with the reference clock. The sampling clock is supplied to a frequency divider  114  and a D/A converter  116 . The frequency divider  114  frequency-divides the sampling clock at a given frequency dividing ratio so that a clock having a frequency approximately equal to that of the reference clock is generated and supplied to the comparator  110 . Hence, a PLL is configured. The D/A converter  116  converts transmission data supplied via a terminal  118  by the above sampling clock into an analog signal, which is then output via a terminal  117 .  
         [0092]    The adder unit  62 , the integral unit  66  and the adder/integral unit  111  have the respective VCO control signals having an approximately identical level. Hence, the control signal level of the integral unit  66  is divided by the resistors R 1  and R 2  with respect to the output (reference) level of the adder/integral unit  111 . The divided voltage is thus applied to the VCO  113 . Thus, it is possible to reduce the capacitor C of the integral unit  66 .  
         [0093]    A further description will be given of reduction in the capacitor C.  
         [0094]    The adder unit  62 , the integral unit  66  and the mixing unit  67  shown in FIG. 13 can equivalently be depicted as shown in FIG. 14A. A current switch  62   a  corresponds to the current switches  83  and  87 , and a voltage source  215  corresponds to the voltage from the adder/integral unit  111 . It will now be assumed that the voltage developed across the capacitor C is denoted as V CO , and the voltage of the connection node between the resistors R 1  and R 2  is denoted as V C . The integral unit  66  and the mixing unit  67  shown in FIG. 14A are equivalently be shown by a capacitor C X  of a voltage V CX . In this case, the following condition stands: 
           C   X   =C· ( R   1   +R   2 )/ R   2 . 
         [0095]    When the current switch  62   a  is turned ON for time dt corresponding to the high-level period of a signal shown in FIG. 15(A) and the capacitor C is charged by current i, an increase dV CO  of the voltage VCO, an increase d VC  of the voltage V C , and an increase dV CX  of the voltage V CX  shown in FIG. 15(B) are expressed as follows: 
         
       dV 
       CO 
       =i·dt/C. 
     
         [0096]    The following is obtained when signal amplitudes in ac formation are considered: 
           dV   C   =dV   CO   ·R   2 /( R   1   +R   2 ) 
         [0097]    and thus 
           dV   C   =i·dt·R   2 /(( R   1   +R   2 )· C )= i·dt/ (( R   1 + R   2 )· C/R   2 )= i·dt/Cx=dV   CX . 
         [0098]    It can be seen from the above that the voltage dividing arrangement using the resistors R 1  and R 2  increases the capacitance of the capacitor C.  
         [0099]    The end of the resistor R 2  is required to be maintained at a constant potential. A transmitter/receiver device equipped with the clock regenerating device has a transmission data output circuit. The level of the VCO control signal output by the adder/integral unit  111  of the PLL substantially to the above constant potential. Hence, the components provided in the system from the comparator  110  to the frequency divider  114  are present in the transmitter/receiver device. In other words, part of the circuit shown in FIG. 13 can be configured by using the existing components of the transmitter/receiver device.  
         [0100]    [0100]FIG. 16 is a circuit diagram of the VCO  54 . The VCO control signal generated by the combining unit  60  is applied, via a terminal  120 , to a voltage-to-frequency (V/I) converter  121 . The converter  121  generates a current dependent on the voltage of the VCO control signal. Currents corresponding to the current generated by the converter  121  flow in charge current switches  122  and  123  in current-mirror formation. Each of the current switches  122  and  123  alternately turns ON and OFF in response to an output clock of an SR-type flip-flop  126 , as shown in FIGS.  17 (D) and  17 (E), so that a capacitor C 10  is charged and discharged. Hence, the voltage V C10  of the capacitor C 10  is changed as shown in FIG. 17(A), and is compared with reference voltages V ref1  and V ref2  by comparators  124  and  125 . The SR-type flip-flop  126  is set and reset by the output signals of the comparators  124  and  125 . The output clock of the SR-type flip-flop  126  shown in FIG. 17(B) is supplied to the current switches  122  and  123  and to a D-type flip-flop  127 , which frequency-divides the output signal of the flip-flop  126  at a frequency dividing ratio of 1/2. Hence, a regenerated clock having a duty ratio of 50% as shown in FIG. 17(C) is available at a terminal  128 .  
         [0101]    A detailed description will now be given of the current switches  122  and  123 . The output current of the V/I converter  121  flows N-channel MOS transistors NT 11  and NT 12  of the current switch  123 . The transistor NT 11  forms current-mirror circuits in cooperation with N-channel MOS transistors NT 13  and NT 16 . The transistor NT 12  forms current-mirror circuits in cooperation with N-channel MOS transistors NT 15  and NT 17 . An N-channel MOS transistor NT 14  is cascaded between the transistors NT 13  and NT 14 , and has a gate supplied with the output clock of the flip-flop  126 .  
         [0102]    The drain currents of the transistors NT 16  and NT 17  are supplied via P-channel MOS transistors PT 11  and PT 12 , which form current-mirror circuits in cooperation with P-channel MOS transistors PT 13  and PT 15 . A P-channel MOS transistor PT 14  is cascaded between the transistors PT 13  and PT 15 , and has a gate supplied with the output clock of the flip-flop  126 . The drains of the transistors PT 13  and NT 13  are connected to the capacitor C 10 .  
         [0103]    The fine current-mirror operation can be realized by the cascaded current-mirror current sources (the current switch  122  includes the transistors PT 11 , PT 13 , PT 12 , PT 15 , and the current switch  123  includes the transistors NT 11 , NT 13 , NT 12  and NT 15 ). In this case, the gate-source voltages of the transistors PT 13  and NT 15  primarily determines the magnitude of the output current, and the transistors PT 15  and NT 13  function additionally. Hence, the switching transistors PT 14  and NT 14  can be provided on the source sides of the transistors PT 13  and NT 15 . Hence, switching noise can be absorbed to a certain extent due to the presence of the transistors PT 15  and NT 13  provided between the switching transistors PT 14  and NT 14  and the capacitor C 10 .  
         [0104]    A description will now be given, with reference to FIGS. 18 and 19, of a variation of the current switch using the current-mirror current source having a pair of transistors.  
         [0105]    [0105]FIG. 18 is a circuit diagram of a variation of the current switch  122 . Referring to FIG. 18, P-channel MOS transistors PT 22  and PT 24  form a current-mirror circuit. The source of the transistor PT 24  is connected to the drain of a P-channel MOS transistor PT 23 . The gate of the transistor PT 23  is supplied with a clock for switching via a terminal  130 . The drain of the transistor PT 24  is coupled to the capacitor via a terminal  131 . A P-channel MOS transistor PT 21  having a drain connected to the source of the transistor PT 22  is provided to be balanced on the transistor PT 23 , and is always ON. Hence, the drain-source voltages of the transistors forming the current-mirror circuit are made substantially identical on both input and output sides thereof.  
         [0106]    [0106]FIG. 19 shows a variation of the current switch  122 . Referring to FIG. 19, N-channel MOS transistors NT 21  and NT 23  form a current-mirror circuit. The source of the transistor NT 23  is connected to the drain of an N-channel MOS transistor NT 24 . The gate of the transistor NT 24  is supplied with a clock for switching via a terminal  132 . The drain of the transistor NT 23  is coupled to the capacitor C 10  via a terminal  133 . An N-channel MOS transistor NT 22  having a drain connected to the source of the transistor NT 21  is used to be balanced on the transistor NT 24 , and is always ON. Hence, the drain-source voltages of the transistors forming the current-mirror circuit are made substantially identical on both input and output sides thereof.  
         [0107]    The current switches shown in FIGS. 18 and 19 may be used to form the current switches  83 ,  84 ,  87 ,  88 ,  89  and  90 .  
         [0108]    Each of the adder unit  62  and the integral unit  66  of the combining unit  60  has a charge pump type low-pass filter having the principle shown in FIG. 20A. Alternatively, it is possible to use an inverted amplifier  220  having feedback elements of the resistor R 10  and the capacitor C 10 .  
         [0109]    The present invention is not limited to the specifically disclosed embodiments, and variations and modifications may be made without departing from the scope of the invention.