Abstract:
A single-antenna short-range radar transceiver emits 24 GHz RF sinewave packets and samples echoes with strobed timing such that the illusion of wave propagation at the speed of sound is observed, thereby forming an ultrasound mimicking radar (UMR). A 12 GHz frequency-doubled transmit oscillator is pulsed a first time to transmit a 24 GHz harmonic burst and pulsed a second time to produce a 12 GHz local oscillator burst for a sub-harmonically pumped, coherently integrating sample-hold receiver (homodyne operation). The time between the first and second oscillator bursts is swept to form an expanded-time replica of echo bursts at the receiver output. A random phase RF marker pulse is interleaved with the coherent phase transmitted RF to aid in spectrum assessment of the radar&#39;s nearly undetectable emissions. The low-cost transceiver can be used for automotive backup and collision warning, precision radar rangefinding for fluid level sensing and robotics, precision radiolocation, wideband communications, and time-resolved holographic imaging.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to wide bandwidth pulsed microwave transmitters and receivers, and more particularly to short-range, sub-nanosecond pulse, phase-coherent K-band radars. 
     2. Description of Related Art 
     Range measurement of close-range targets is of great interest to a number of industries. Automotive backup warning radar, fluid level sensing in tanks and vats, material level sensing in silos, safety systems, home “do-it-yourself” projects, and aids to the blind are but a few of the applications for non-contact short-range measurement. 
     Ultrasound is a ranging technology that is both simple and inexpensive. Unfortunately, it is of limited accuracy since the speed of sound varies 10% over outdoor temperatures. Accuracy is of central importance in tank level measurement and construction applications, where accuracies of 1% to 0.01% are needed. In addition to limited accuracy, ultrasound is susceptible to extraneous acoustic noise, and water or dirt overcoatings on its transducers can disable it. In spite of these limitations, ultrasound has been a popular ranging technology due to its simplicity and its ability to form a narrow beam with a small transducer. 
     Radar rangefinders are environmentally rugged: the speed of light (at which radar waves travel) does not vary with temperature (for all practical purposes), and radar waves propagate freely through wood walls, gypsum walls and plastic panels, even with an overcoating of water, ice, snow or dirt. Pulse-echo radars operating in the 24 GHz band have a wavelength of 12.5 mm, which is almost exactly the same wavelength as 24 kHz ultrasound. Since antenna beamwidth is determined by the wavelength to antenna aperture ratio, radar and ultrasound will have comparably narrow beamwidths with the same antenna/transducer footprint. 
     An ultrasonic rangefinder may typically transmit a burst of 12 sinusoidal cycles of acoustic energy with a corresponding pulse width that defines the two-object resolution of the system. Of course, its incremental resolution is not a function of emitted pulse width, but that of the timing system. A 24 GHz radar with the same two-object resolution as the 12-cycle ultrasound system needs to transmit a 12 cycle, 0.5-nanosecond sinusoidal burst at 24 GHz, since the wavelengths are comparable. Clearly, the radar needs to have a wide bandwidth, on the order of 2 GHz. 
     With the exception of copending application Ser. No. 09/238,455 by McEwan filed Jan. 28, 1999now U.S. Pat. No. 6,191,724, prior art pulse-echo radars do not exhibit the combination of 1) K-band RF operation, e.g., 24 GHz, 2) sub-nanosecond RF pulse width, 3) extreme phase coherence (&lt;10-picoseconds for the entire transmit-receive system, 4) expanded time output with ultrasonic parameters, 5) simple assembly with low cost surface mount technology (SMT) components, and 6) commercially appealing size and cost. 
     SUMMARY OF THE INVENTION 
     The present invention is a short-range radar transceiver (transmitter-receiver) that uses the same pulsed-RF oscillator as a transmit oscillator and as a swept-in-time pulsed receive local oscillator. This dual function use of one oscillator eliminates the need for two microwave oscillators and facilitates operation with only one antenna for both transmit and receive functions. Further, it assures optimal operation since there are no longer two oscillators that can go out of tune with each other (in a two oscillator system, both oscillators must be tuned to the same frequency). 
     In operation, a short sinusoidal RF burst is transmitted to and reflected from a target. Shortly after transmission, the same RF oscillator used to generate the transmit pulse is re-triggered to produce a local oscillator pulse (homodyne operation), which gates a sample-hold circuit in the receiver to produce a voltage sample. This process is repeated at a several megaHertz rate. With each successive repetition, another sample is taken and integrated with the previous sample to reduce the noise level. Also, each successive local oscillator pulse is delayed slightly from the previous pulse such that after about 10 milliseconds, the successive delay increments add up to a complete sweep or scan of perhaps 100-nanoseconds, or about 15 meters in range. After each scan, the local oscillator delay is reset to a minimum and the next scan cycle begins. 
     The incremental scan technique produces a sampled voltage waveform on a millisecond scale that is a near replica of the RF waveform on a nanosecond scale. This equivalent time effect is effectively a dimensionless time expansion factor. If the expansion factor is set to 1-million, 24 GHz sinewaves are output from the system as 24 kHz sinewaves. Accordingly, the radar output can be made to appear like an ultrasonic ranging system. In addition to having the same frequency, e.g., 24 kHz, a 24 GHz radar actually has the same wavelength as a 24 kHz ultrasonic system. In addition, the range vs. round-trip time is the same (in equivalent time for the radar, of course). FIGS. 3 a  and  3   b  compare the radar of the present invention to an ultrasound rangefinder. The responses are similar, even though, one is electromagnetic and the other is acoustic. The present invention can be dubbed ultrasound mimicking radar (UMR). 
     Precision timing circuits are required for accurate expansion factors. Timing circuits having scale factor accuracies on the order of several tens of picoseconds or better can be realized with a Delay Locked Loop (DLL) such as a “Precision Digital Pulse Phase Generator” as disclosed by McEwan in U.S. Pat. No 5,563,605, or in copending application, “Phase-Comparator-Less Delay Locked Loop” Ser. No. 09/084,541, filed May 26, 1998, now U.S. Pat. No. 6,055,287, by McEwan. Alternatively, dual crystal clocks, one for transmit and one for receive, can be employed, where the receive clock is locked to a small offset frequency from the transmit clock, such as 100 Hz, thereby causing a steady phase slip of one complete clock cycle 100 times per second. In the process, the receive sampler timing smoothly sweeps across one complete pulse repetition interval (e.g., PRI=100 ns for a 10 MHz clock) every 10 ms in equivalent time. These dual oscillator timing circuits are described in copending applications “Self Locking Dual Frequency Clock System,” Ser. No. 09/282,947 filed Apr. 1, 1999, and “Precision Radar Time Base Using Harmonically Related Offset Oscillators,” Ser. No. 09/285,220, filed Apr. 1, 1999, now U.S. Pat. No. 6,072,427, both to McEwan. 
     The emission spectrum from a short-pulsed RF oscillator is very broad (often greater than 1 GHz) and appears very low in amplitude on a spectrum analyzer of limited bandwidth, e.g., 1 MHz bandwidth, as preferred in FCC tests. Consequently, a narrowband, incoherent RF marker pulse is interleaved with the short coherent RF pulses used for ranging to produce a very visible spectrum with an identifiable peak, i.e., carrier frequency. However, the marker pulse creates spurious echoes. Accordingly, the marker pulse is randomized in phase so its echoes average to zero in the receiver. At the same time, the desired ranging pulses phase-coherently integrate from pulse to pulse into a clean signal. 
     The present invention is a precision radar rangefinder that can be used in radars for many applications, e.g., tank level measurement, including 0.01% accurate custody transfer measurement; industrial and robotic controls; vehicle backup warning and collision radars; and general rangefinding applications. Since the present invention is phase coherent, microwave holograms can be formed using techniques known in the art, where the customary holographic reference beam is conveniently replaced by the internal phase coherent timing of the present invention. 
     A primary object of the present invention is to provide a precision, low cost radar ranging system with a narrow beamwidth using a single, small antenna and with an ultrasound-like output. 
     Yet another object of the present invention is to provide a wideband radar ranging system with a measurable center frequency. 
     Still another object of the present invention is to provide a radar having a single transmit/receive oscillator and a harmonic sampler that can be fabricated with common surface mount components. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a block diagram showing the single-antenna short-pulse microwave radar transceiver of the present invention. 
     FIG. 2 is a schematic diagram of the single-antenna short-pulse microwave radar transceiver of the present invention. 
     FIG. 3 a  plots the recorded baseband video output from a single antenna radar of FIG.  1 . 
     FIG. 3 b  plots the recorded baseband video output from an ultrasonic rangefinder (prior art). 
     FIG. 4 a  plots the recorded RF emission in the time domain. 
     FIG. 4 b  plots the recorded emission spectrum with and without a marker pulse. 
     FIG. 4 c  plots the recorded ranging pulse (T) and its 256-repetition average (T int ). 
     FIG. 4 d  plots the recorded marker pulse (M) and its 256-repetition average (M int ). 
     FIG. 5 is a timing diagram showing the relation between the clocks and the RF bursts. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     A detailed description of the present invention is provided below with reference to the figures. While illustrative component values and circuit parameters are given, other embodiments can be constructed with other component values and circuit parameters. All U.S. Patents and copending U.S. applications cited herein are herein incorporated by reference. 
     FIG. 1 is a block diagram of a microwave transceiver  10  of the present invention. A harmonic oscillator  12  receives clock pulses from transmit (TX) clock  14  via pulse driver  18  and produces RF burst pulses (transmit pulses T) at the transmit antenna  16 . TX clock  14  typically produces a 1-10 MHz squarewave that is passed through pulse driver  18  to form ˜1 ns wide pulses with rise and fall times on the order of 100 ps. Thus, the drive pulses are clock pulses with very fast rise and fall times. Therefore, the TX clock  14  and pulse driver  18  may together be viewed as a clock signal generator. These short pulses bias-on the harmonic oscillator  12 , which is designed to start and stop oscillating very rapidly as a function of applied bias. The oscillations of the T pulses are phase coherent with the drive pulses, i.e., the phase of the RF sinusoids relative to the drive pulse remains constant, i.e., coherent, each time the oscillator is started—there is no significant clock-to-RF jitter. However, as will be discussed below with reference to the marker generator  21 , separate marker pulses M may have a random phase relative to the clock. 
     A high degree of phase coherence for the T pulse can only be obtained with a very fast risetime drive pulse that shock excites the oscillator into oscillation. Accordingly, the pulse driver of the present invention has transition times of about 100 ps to ensure coherent oscillator startup. 
     Harmonic oscillator  12  typically operates at a fundamental frequency of 12.05 GHz with a second harmonic at 24.1 GHz. A frequency of 24.1 GHz or thereabouts is preferred since commercial and consumer devices such as radar rangefinders can operate in the 24.0-24.25 GHz band without a license. The transmitted RF bursts (T) are typically 12 cycles long at a carrier frequency of 24.1 GHz. 
     In a less preferred mode, a fundamental frequency oscillator  20  operating at 24.1 GHz can be used instead of a harmonic oscillator  12 . A fundamental mode oscillator has a higher output amplitude but is far more difficult to implement in surface mount technology (SMT) since a quarter-wavelength at 24 GHz is 3 mm, about the same length as a SMT transistor package (including leads) that would be used for the oscillator. Obtaining oscillations in a SMT package at 24 GHz is nearly impossible. 
     A receive (RX) clock  22  typically produces a 1-10 MHz squarewave that is passed through pulse driver  18  to form ˜1 ns wide pulses with rise and fall times below 100 ps. These short pulses bias-on the harmonic oscillator  12  in a similar fashion to the drive pulses triggered by the TX clock to form 0.5 ns wide gate pulses G. The G pulses gate the harmonic sampler  30  at typical frequency of 12 GHz via path  23  to sampler  30 . Alternatively, a fundamental oscillator  20  is employed and gates sampler  30  at 24 GHz via path  25 . Sampler  30  develops a detected signal, representing the coherent integration of multiple gatings of sampler  30 , which is amplified by a low frequency amplifier  31  and filtered in bandpass filter  32  to produce a baseband video signal. Sampler  30  samples reflected signals R received by antenna  16 . 
     The RX clock  22  is typically swept in time relative to the TX clock  14 . A swept delay RX clock can be realized by the timing control unit  34  alone (acting as a delay element with controlled delay), when connected between TX clock  14  and pulse driver  18 , with the offset frequency RX clock  22  omitted (dashed lines) and timing control unit  34  providing the RX clock. A means to accomplish precision swept timing has been described in co-pending application “Phase-Comparator-Less Delay Locked Loop,” Ser. No. 09/084,541, filed May 26, 1998, now U.S. Pat. No. 6,055,287, by McEwan, which exhibits 0.01% linearity across a 10-meter measurement range. 
     Alternatively, swept timing can be realized by operating an RX clock  22  at a slightly lower, or offset, frequency than TX clock  14 . Typically, the TX clock is a first quartz crystal oscillator, and the RX clock is a second quartz crystal oscillator that is phase locked to a typical offset of 100 Hz from the TX clock, such that the phase of the RX clock slips one full pulse repetition interval (PRI) every 10 ms. In this case, timing control unit  34  performs a phase-lock function between the TX and RX clocks and its timing output is not present, i.e., it is not directly connected to pulse driver  18  as indicated by the “X”  35 . 
     With either type of swept timing, an equivalent time (ET) replica of the transmitted RF signal radiated from antenna  16  and received by the same antenna, or alternatively by antenna  28 , (or radiated from a remote transmitter in the case of a radiolocation or communication system) appears at the harmonic sampler  30  output. When a separate receive antenna  28  is used, no path exists between antennas  16  and  28 , as indicated by “X”  37 . The ET replica resembles the RF signal, except it occurs on a slow time scale such as 10 ms. FIG. 3 a  provides an example of an ET replica signal on a 2-millisecond/div scale. In the typical case where the harmonic oscillator gate pulse G contains more than one RF cycle, the ET output is broadened somewhat by the convolution of the harmonic oscillator transmit pulse T and the harmonic oscillator gate pulse G. 
     The TX clock  14  can be modulated in frequency or phase, or with ON-OFF gating by optional modulator  33  for various purposes known in the art, such as spreading the spectral lines generated by harmonic oscillator  12  to reduce interference to other spectrum users. The modulation may be coded to distinguish the desired received signal from that of others. Modulator  36  is connected to the RX clock  22  or timing circuit  34  to demodulate coded signals from a remote transmitter using synchronization means know in the art of spread spectrum radio technology. 
     A marker generator  21  is triggered by pulses from the pulse driver to form marker pulses M which are much wider than the T or G pulses. Due to the width of the M pulses, the radiated spectrum becomes relatively narrow, since the emission spectrum is roughly related by 1/PW, where PW is the width of the emitted pulses. One purpose of the narrow marker pulse spectrum is to aid in identifying the RF carrier frequency and spectral width of the transmitted pulses. 
     FIG. 2 is a detailed schematic of the pulse driver  18 , harmonic oscillator  12 , antenna  16 , sampler  30 , amplifier  31  and bandpass filter  32  of the present invention. The TX clock signal is coupled through capacitor  40  and switching transistor  42  (which forms pulse driver  18 ) to provide a transmit pulse with a fast leading edge. Capacitor  40  and potentiometer  46  provide an adjustable time constant that defines the width of transmit pulse T, generally from 0.5 to 5 ns. 
     The particular setting of the pulse width depends on the RF parameters required of the RF system. Very short RF pulses are needed for high spatial resolution or for high data rate systems, whereas wide pulses allow for a narrower bandpass filter  32  at the video output and thus better signal to noise ratio. Also, the RF pulse width may be limited by regulatory constraints where a very short pulse may have excessive bandwidth. 
     Harmonic oscillator  12  is comprised of a GaAsFET  54 , and several microstrip resonators labeled L 1 , L 2 , L 3  connected to the source and drain of GaAsFET  54 ; antenna  16  is also connected to the drain. The microstrip resonators L 1 , L 2 , and L 3  are all approximately one-quarter wavelength long at the fundamental frequency of oscillation, which is 12.05 GHz for a 24.1 GHz radiated frequency. Antenna  16  is comprised of two feeds  16   a ,  16   b  which are quarter-wave resonant at the second harmonic, or 24.1 GHz, and combined into a single waveguide  77  to form a single antenna  16  when viewed from outside waveguide  77 . Antenna element  16   a  resonates at 24 GHz and provides a low impedance to the drain of the GaAsFET for maximum harmonic extraction. Experiments show that second harmonic generation at 24.1 GHz is only 4 dB lower than the same transistor operating in the fundamental mode at a frequency of 24.1 GHz. 
     One of the microstrip resonators, either L 2  or L 3 , can be adjusted in length or may have a short wire or metal tab (not shown in FIG. 2) attached to it to fine-tune the oscillator frequency. Alternatively, a metal screw  55  can be brought close to one of the microstrips for fine-tuning. 
     The entire oscillator circuit  12  is placed inside a waveguide  77  that operates beyond cutoff at 12 GHz to substantially reduce unwanted radiation at the 12 GHz fundamental frequency. Thus, waveguide  77  is somewhat greater than one-quarter wavelength high (so the quarter-wave antenna fits inside) and one-half wavelength wide at 24 GHz. A horn  78  is coupled to the waveguide to provide gain and reduce sidelobe radiation. In one prototype, the horn flares from 4×6 mm to 20×50 mm and provides about 18 dB gain at 24 GHz. 
     Harmonic oscillator  12  operates at 12 GHz (but produces a transmit signal at 24 GHz) and has three key advantages over a fundamental mode oscillator  20  operating at 24 GHz. First, all the microstrip dimensions are twice as long. This is critical, since the microstrip dimensions at 24 GHz become smaller than the SMT transistor leads. Second, the harmonic oscillator can start up very rapidly since the GaAsFET has much higher gain at 12 GHz than at 24 GHz. This is particularly true when package parasitics are factored in. There appears to be a “wall” at about 22 GHz, beyond which SMT devices will not oscillate without special layout techniques. 
     The third advantage to a harmonic oscillator is that the startup phase of oscillation is very independent of external RF at double frequency, i.e., received signals at 24 GHz. 
     Experiments show that the slightest RF present at 24 GHz will alter both the startup phase and the oscillation envelope risetime of a 24 GHz oscillator. Consequently, vitally needed phase coherence is lost, and when the oscillator is used as a local oscillator, the external-RF modulated local oscillator envelope causes extremely high noise in the detector. 
     GaAsFET transistor  54  is a NEC type NE42484A, silicon bipolar transistor  42  is an NEC type NE68533, and inverter  50  (described below) is type 74AC04. 
     The diodes (D 1 -D 4 ) of sampler  30  receive 24 GHz echoes from receive monopole  16   b  plus 12 GHz oscillations from oscillator  12 , coupled from resonator L 2  via a proximity coupling path indicated by dashed line  53 . Thus the input to sampler  30  is comprised of an algebraic sum of 24 GHz echoes and 12 GHz oscillator G pulses (as well as T pulses, which are generally excluded from the swept timing window). 
     The back-to-back connection of detector diodes D 1 -D 2  and D 3 -D 4  results in conduction on each half cycle of the 12 GHz oscillation provided by oscillator  12 . The net effect is the same as a single diode conducting on every full cycle of a 24 GHz local oscillator. Accordingly, the sampler  30  operates on the second harmonic of oscillator  12 . Experiments show that this frequency doubling harmonic sampler has nearly the same sensitivity as a non-harmonic 24 GHz local oscillator and sampler. In addition to harmonic sampling, a key advantage to the back-to-back diode connection is rejection of local oscillator noise, since the back-to-back connection develops zero offset bias (in principle), and consequently there can be little noise due to the local oscillator. 
     Quarter wave microstrips L 4  provide 24 GHz RF shorts at the video output end of the sampler to prevent spurious RF leakage to the amplifier  31 . These RF shorts also enhance the detection efficiency of diodes D 1 -D 4 . Amplifier  31  amplitude-scales and impedance-buffers the detected signal. The signal is then passed through a bandpass filter  32  to limit noise and, in some cases, to provide Doppler filtering for velocity discrimination. 
     Sampler diodes D 1 -D 4  are connected as series pairs (D 1 -D 2  and D 3 -D 4 ) and then as back-to-back (or anti-parallel) pairs. The diodes are connected in series to cut the terminal capacitance in half. Yet the terminal inductance of two leads in a series connection remains about the same as two leads for a single diode. This assumes the diodes are on a common substrate and packaged in a single SMT package. The inductance between the diodes connected in series is limited to a small on-chip inductance and is negligible compared to the lead inductance of the package. Experiments show that a single 0.25 pF Schottky diode in a small SMT package has a detector bandwidth of about 10 GHz, whereas a series pair (e.g., D 1 -D 2  or D 3 -D 4 ) has a bandwidth of greater than 20 GHz. Schottky sampling diodes D 1 -D 4  are HP type HSMS-286C and the op amp  86  of low frequency amp  31  is a TI type TLO71. 
     Ballast resistors  56  and  57  are connected to the diode outputs to provide a defined detection current though the detector diodes. A steady positive detection voltage is developed across resistor  57  and a steady negative voltage is developed across resistor  56  as the result of repetitive conduction and peak holding action of the diodes in conjunction with several pico-Farads of stray capacitance. Capacitor  58  provides an effective short-circuit at all detected, i.e., video frequencies of interest, so the combined signal outputs of the two branches of diode  30  are conducted to amplifier  31 . Out-of-phase noise appearing across resistors  56 ,  57  is shorted-out by capacitor  58 , resulting in an effective cancellation of noise from oscillator  12 . At the same time, ballast resistors  56 ,  57  help maintain a balanced conduction of the two branches of sampler  30  to help ensure a high level of oscillator noise rejection (better balance leads to greater cancellation of the positive and negative noise components appearing at the output of sampler  30 ). 
     A receive (RX) clock signal is coupled to driver  18  via a gating path (shown as G-path in FIG. 2) which is similar to the transmit path (shown as T-path in FIG.  2 ), i.e., through capacitor  41  to transistor  42 . In the gating path, the timing of the receive clock is swept in phase relative to the transmit clock to provide a range scan. Each positive edge of either the TX or the RX clock will switch-ON transistor  42  for about 1-nanosecond, i.e., for a duration set by capacitors  40  and  41  for the T pulse and the G pulse respectively, and in conjunction with potentiometer  46 . 
     A marker pulse is produced through a marker path (shown as M-path in FIG. 2) comprised of a timing network  51  connected to TX clock, a logic inverter  50  to set operation on the negative-going edge of the TX clock, a shaping network  52 , potentiometer  46 , and then through switching transistor  42  to oscillator  12 . On each negative edge of the TX clock, a 60 ns wide pulse is coupled through path M to produce a 60 ns wide oscillator pulse which radiates from horn  78  to create a relatively narrow marker spectrum relative to the broad low-amplitude spectrum radiated by the 1 ns transmit pulse T, thereby marking the center frequency of the emissions from transceiver  10 . Without the marker pulse, the emission spectrum is so broad and flat that there is no discernable center frequency. 
     Shaping network  52  slows the transition times of the pulses coupled to driver transistor  42 , and in turn it slows the transition times of the driver pulses coupled to oscillator  12 . With sufficiently slow drive pulses, typically with transition times of 0.5 ns or slower, oscillator  12  will start oscillating with a random phase relative to the drive pulse transitions. Thus, on each repetition of the TX clock, a random-phase marker pulse is emitted (random with respect to the negative edges of the TX clock, and random to the positive edges of both the TX and RX clocks). Since sampler  30  integrates multiple repetitions of gated echoes, echoes produced by the marker pulses are integrated to zero, since random-phase pulses average to zero after many pulse integrations. 
     Alternatively, marker path M may be coupled from the gating path G, i.e., network  51  is connected to RX clock such that marker pulses are generated with each negative transition of the RX clock. In this case, the marker pulses may be coherent with respect to either or both the TX and RX clock, since echoes from the marker pulse are stationary in time relative to the receive clock, i.e., there is no time sweep relative to the RX clock, and so the output from sampler  30  due to the marker pulses is at zero frequency, i.e., DC. Accordingly, bandpass filter  32  rejects this DC component such that the presence of the marker pulses has no effect on the normal operation of transceiver  10  using the T and G pulses. Waveform  44  shows the relation between pulses T, G and M. 
     FIG. 3 a  plots actual data for the video output of radar transceiver  10  with its timing set to produce an equivalent time output having characteristics matching an ultrasonic rangefinder. The time scale is set by adjusting the sweep rate of the RX clock to produce an equivalent time scale of 0.15 meters/millisecond, exactly the same as a ranging transceiver based on ultrasound. Thus, the transceiver is an ultrasound mimicking radar (UMR). In addition to matching the time scale with ultrasound, the transceiver output of FIG. 3 a  is a burst having a frequency of 24 kHz, exactly as though it were a 24 kHz ultrasound burst, and not a 24 GHz radar burst as it is in reality. Furthermore, the radar wavelength at 24 GHz is nearly an exact match for an ultrasonic system operating at 24 kHz, so the radar antenna can have a matching footprint to an ultrasonic transducer of the same beamwidth. 
     FIG. 3 b  plots actual data from a 40 kHz ultrasound system using commercial 40 kHz transmit and receive transducers and ranging to the same metal target plate at the same 3-meter range as the UMR of FIG. 3 a . As a comparison, the UMR output of the present invention is seen to be essentially the same as the output of an ultrasonic rangefinder. Thus, ultrasonic transducers can be exchanged with UMR while retaining the same video processing, e.g., automatic gain control, threshold detection and range counting. 
     FIG. 4 a  plots actual data for the radiated signal from antenna  16 . A short transmit pulse T is radiated with a pulse width of 500 ps followed by a 60 ns wide marker pulse M. The 24 GHz sinusoids cannot be seen because they are too closely spaced to be resolved in the plot—the T pulse alone contains about 12 cycles. Also, the thick trace between the T and M pulse arises from noise in the oscilloscope combined with the plot being set for peak-envelope mode. 
     FIG. 4 b  shows the spectrum of the waveform of FIG. 4 a , as the T+M trace. The 24 GHz band, as defined by the FCC, has been overlaid to show that the T+M pulse fits in the band and any signal that falls outside the 24 GHz band is below the spurious limit. Trace T shows the spectrum of the T pulse alone, with the marker pulse M deleted. It is barely above the noise floor, even though the antenna spacing from antenna  16  to the spectrum analyzer antenna used to take this data was only 3 cm. While the spectrum from T is barely detectable, it can create problems for regulatory agencies interested in measuring center frequency and spectral width. 
     FIG. 4 c  plots actual data for a coherent phase 24 GHz 500-picosecond wide transmit burst T, as recorded on a Tektronix sampling oscilloscope (upper trace) and a 256 trace integration, or waveform average, of 256 iterations of the transmitted 24 GHz RF pulse T (lower trace). The reduction in noise with coherent integration is very pronounced. 
     FIG. 4 d  plots actual data under the same conditions as in FIG. 4 c  for a random phase 24 GHz marker pulse M. The pulse structure seen in the upper trace is random since the phase is random. The lower trace shows the impact of 256 waveform integrations: the M pulse is effectively averaged to zero. This is in contrast to the clean pulse created in FIG. 4 c  by the integration of the coherent phase pulses. Accordingly, spurious echoes of the M pulse integrate to zero and do not create an interference signal, i.e., a ghost reflection. The transceiver of FIG. 1 performs integration partially or wholly in the sampler  30  and may further integrate the sampled signal by means of the band-limiting, or integrating, action of bandpass filter  32 . 
     FIG. 5 illustrates a timing diagram for the transceiver of FIG.  1 . One cycle of the TX clock is shown in the top trace as a reference waveform. At each positive transition of the TX clock a T pulse is emitted with a coherent, i.e., consistent, phase relative to the TX clock. At each negative transition of the TX clock an M pulse is emitted with a random phase relative to the TX clock. These pulses are shown as solid blocks to indicate a large number of sinusoids within each block. The next lower trace, RX RF, depicts the echoes from the T and M pulse emissions for two targets at ranges R 1  and R 2 . The GATE RF trace shows the timing relation for gate pulses G, which are swept in time relative to the TX clock, from G min  to G max , or the minimum and maximum ranges in the range sweep. Whenever the G pulse is coincident with an echo pulse at R 1 , sampler  30  will produce an output for that point in the sweep of the G pulse. Since the reflection of a T pulse from a target at R 2  ( 60 ) is beyond the maximum sweep at G max , it will not be detected. However, a reflection of an M pulse from a target at R 2  ( 62 ) falls into the next clock cycle and into the sweep range of G. It appears as a range ambiguity, i.e., it looks like an in-range target. Since the phase of the M pulse is random, so too is the phase of its echo at R 2  ( 62 ). Thus, coherent integration of this echo by sampler  30  and bandpass filter  32  will average this spurious reflection to near-zero. 
     The bottom trace in FIG. 5 shows an optional timing arrangement for the marker pulse M, wherein the M pulse is initiated by the negative transition of the receive clock with coherent phase or incoherent phase relative to the RX clock. This negative transition can be swept in time just as the G pulse is swept, such that the time between the G and M pulses remains somewhat constant. Consequently, sampler  30  will always sample the same part of the reflected M pulse  62  and produce a DC or near-DC output. Bandpass filter  32  can then filter out these low frequency components. With either timing relation for the M pulse, spurious echoes from the M pulse are minimized. 
     Although the invention has been described with reference to a 24 GHz system because of the high interest in this frequency, the principles of the invention can be applied to other frequencies, e.g., 10.5 GHz and 38 GHz. The techniques using harmonic transmit pulses and sub-harmonic sampling can be similarly applied. 
     Changes and modifications in the specifically described embodiments can be carried out without departing from the scope of the invention which is intended to be limited only by the scope of the appended claims.