Abstract:
A delay circuit comprising a plurality of cascaded saturating circuit elements is provided. The delay circuit may be incorporated in such circuits as modulators and demodulators to provide signal delay.

Description:
BACKGROUND OF THE INVENTION 
     Analog signal processing applications often require the use of signal delay lines. Such delay lines can be categorized generally as either analog or digital in nature. A typical example of an analog delay line is that disclosed by Puckette et al. in U.S. Pat. No. 3,973,138 issued Aug. 3, 1976 which uses a bucket brigade of serially connected capacitors to provide a desired amount of time delay. Such analog delay lines are unfortunately relatively expensive, require the use of analog switches, and tend to suffer from crosstalk problems. In addition, either when used singly or when cascaded, such analog systems invariably reduce the bandwidth of the signal being processed. 
     A typical digital delay line is disclosed by Covington in U.S. Pat. No. 3,760,280 issued Sept. 18, 1973, in which a single analog signal channel is converted to a frequency modulated (FM) signal which in turn is delayed by means of a clocked digital shift register. The resulting delayed digital signal is then demodulated to provided a delayed analog signal. Such a digital delay system overcomes many of the problems of analog delay lines, but the bandwidth problem still remains. Since the digital signal is propagated through the shift register by means of a clock signal, it is necessary to use a very high speed shift register and clock to maintain the overall system bandwidth. Thus, according to conventional sampling signal theory, in order for the delayed output signal to have a 5 megahertz (MHz) information bandwidth with a 0.1% pulse width resolution, the shift register must be clocked at or above 10 gigahertz (GHz) (i.e., 5 MHz×1000×2). 
     Other workers such as Arnstein in U.S. Pat. No. 4,124,820 issued Nov. 7, 1978 have shown digital delay lines which do not make use of a clocked shift register, but instead achieve their desired delay function by applying an FM signal to a plurality of conventional digital gates arranged in cascade along with latch connected logic gates to reconstitute travelling through the delay circuit. Propagation delay is then adjusted by adding external timing capacitance or resistance to compensate for device variations. Although such an asynchronous delay line does not make use of a clock as in Covington, the resulting output signal is still bandwidth limited due to the low bandwidth of the individual digital gates, the use of latches to overcome propagation losses, and the use of resistors and capacitors to adjust the propagation delay. 
     SUMMARY OF THE INVENTION 
     In accordance with the preferred embodiments of the present invention, a delay circuit is presented for providing signal delay. In order to delay an analog signal, the analog signal is first represented by a binary signal with only two voltage states (a logic 1 and a logic 0). For example, this can be done by using the analog signal to pulse width modulate, frequency modulate or phase modulate a carrier wave to produce a modulated signal. The modulated signal can then be coupled to a plurality of cascaded saturating elements. Each saturating element provides an incremental delay. Delay across the delay circuit can be varied by providing a digital switching means which allows selection of the number of circuit elements through which the binary signal travels, by varying the period of delay through each element, or by a combination of the above two methods. After passing through the delay circuit the binary signal can then be restored by demodulation to the initial analog signal, without loss of bandwidth. 
     The delay circuit for providing signal delay may be incorporated in many types of devices. For instance, the delay circuit may be incorporated in a transversal filter to replace more conventional analog delays, in a wideband FM detector to provide phase shift, or in a phased array acoustic imaging system to align signals from array elements. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic diagram of cascaded inverters in accordance with a preferred embodiment of the present invention. 
     FIG. 2 shows a circuit which uses a switch to vary propagation time through the circuit. 
     FIG. 3 shows a circuit which varies propagation delay from an input to an output by means of switches organized in the form of a binary tree. 
     FIG. 4 shows a circuit which varies propagation delay by means of an analog signal. 
     FIGS. 5A-5B show the coupling of two inverters to form a delay stage. 
     FIGS. 5C-5E show the coupling of two inverters to form a delay stage in accordance with preferred embodiments of the present invention. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT 
     Delay of a binary signal may be accomplished by coupling the signal througha series of saturating elements. What is meant by a binary signal is a signal which changes between two voltage levels. The voltage levels, commonly referred to as logic 0 and logic 1 may be, for example, 0 volts and 3 volts. 
     In FIG. 1, a binary signal (Vb 1 ) is applied to an input 12 of a plurality of cascaded saturating circuit elements, in this case inverters 14, 16, 18, 20, 22, 24, 26, 28, and 30. A supply voltage (Vdd), for example 3 volts, is coupled to the system at a node 32. In the preferred embodiment, each of the plurality of cascaded inverters has a propagation (or delay) time t of a few (e.g. 3-20) nanoseconds. At an output 10, the delay of the binary signal is equal to the number of inverters multiplied by the propagation time through each inverter. In the circuit of FIG. 1, this equals 9t. 
     In FIG. 2, a binary signal (Vb 2 ) is coupled at an input 46 to a plurality of cascaded inverters 34, 36, and 38. An output 42 is coupled toa node 50 or a node 48. A single-pole-double-throw switch 44 or its logic equivalent toggles output 42 between node 50 and node 48, and thus varies the duration of delay from input 46 to output 42. For example, if each of the cascaded inverters has a propagation time t, the delay when output 42 is coupled to node 48 is t and the delay when output 42 is coupled to node50 is 3t. 
     In FIG. 3, an example is shown of a means to vary delay propagation by digital switching utilizing a binary tree. A binary signal is coupled at an input 320 to a plurality of delay elements 321, 322, 323, 324, 325, 326, and 327, typically groups of cascaded inverters as in FIGS. 1 and 2. Typically, when fabricated on a single integrated circuit, delay time T will be the same for each delay element group 321-327. A series of logic switches 331, 332, 333, 334, 341, 342, and 351, which are typically single-pole-double-throw switches or their logic equivalent, are arranged in the form of a binary propagation time through the entire circuit. Depending upon the position of switches 331-334, 341-342, and 351, the range of propagation delay from input 320 to an output 360 is 0T to 7T. Inthe circuit of FIG. 3 there are 3 levels of switches: switch 351 forms the first level; switches 341 and 342 form the second level; and switches 331,332, 333, and 334 form the third order level. The switches at each level for convenience, may be switched together. For instance, as shown in FIG. 3, third level switches 331, 332, 333, and 334 are each selecting a pole labeled &#34;1&#34;. 
     In the embodiment shown in FIG. 3 each level of switches represents a bit in a binary number. Level 1 is the most significant bit, level 2 is the next most significant bit, and level 3 is the least significant bit, e.g.,in FIG. 3 level 1 switch 351 is set at &#34;1&#34;, level 2 switches 341-342 are set at &#34;0&#34;, and level 3 switches 331-334 are set at &#34;1&#34;, so that the current delay is 101 base  2 times T, that is 5T. 
     In FIG. 4, an alternative method to vary delay is illustrated. A binary signal (Vb 4 ) is coupled at an input 72 to a plurality of saturating elements, 64, 66, 68, and 70, typically cascaded inverters, as shown. Variable resistors 74, 76, 78, and 80, for example depletion-type metal-oxide-silicon field effect transistors (MOSFETS), are controlled by a voltage (Vc 4 ) applied to an input 62 and function as variable current sources. Although delay time t varies as Vc 4  varies, typically, when Vc 4  is held constant, delay time t will be the same for each saturating element 64, 66, 68, and 70, when saturating elements 64, 66, 68, and 70 are fabricated on a single integrated circuit. Delay from input 72 to output 60, therefore, is always 4t. When Vc 4  is decreased, resistance across each of variable resistors 74, 76, 78, and 80increases, decreasing current through variable resistors 74, 76, 78, and 80and thereby increasing the propagation (delay) time t of each of the saturating elements 64, 66, 68, and 70. Correspondingly, as Vc 4  is decreased, propagation time t decreases. Therefore, varying Vc 4  varies the delay from input 72 to output 60. 
     FIGS. 5A-5E illustrate the coupling of two inverters on an integrated circuit. FIG. 5A shows a circuit where an inverter 607, comprising an enhancement-type MOSFET 603 and a depletion-type MOSFET 601, is coupled asshown to an inverter 608, comprising an enhancement-type MOSFET 604 and a depletion-type MOSFET 602. MOSFETs 601 and 602 act as load resistances forinverters 607 and 608. An input node 609 is coupled to an output node of a prior inverter. An output node 606 is coupled to an input of a subsequent inverter. As can be seen from FIG. 5A, signal Vo on output 606 is the sameas a voltage Vi on input 609 after a propagation delay through inverter 607and a propagation dely through inverter 608. On this circuit experimental results showed a ratio of rise to fall time to be at least 3:1. What is meant by rise time is the time it takes for signal Vo to rise from logic 0to logic 1 after signal Vo starts to rise from logic 0 to logic 1. What is meant by fall time is the time it takes for signal Vo to fall from logic 1to logic 0 after signal Vo starts its fall from logic 1 to logic 0. The asymmetry in rise to fall times makes it difficult to propagate high frequency pulse trains through a string of inverters, coupled as in the circuit in FIG. 5A, without distortion of the timing between pulses. 
     In FIG. 5B an inverter 631, comprising an enhancement-type MOSFET 623 and adepletion-type MOSFET 621, is coupled as shown to an inverter 632, comprising an enhancement-type MOSFET 624 and a depletion-type MOSFET 622.MOSFETs 621 and 622 act as variable load resistances. An input node 629 andan input node 627 are coupled to output nodes of a prior inverter. An output node 626 and an output node 634 are coupled to inputs of a subsequent inverter. As shown in FIG. 5B, input 629 is coupled to MOSFET 623, and is also coupled to a gate 628 of MOSFET 622. Similarly a node 625of inverter 631 is coupled to MOSFET 624 at a gate 630, and is also coupledto output node 626. The rise to fall ratio from input 629 to output 634 forthe circuit of FIG. 5B was found to be about 1.5:1. This characteristic makes the circuit in FIG. 5B better qualified than the circuit in FIG. 5A to propagate high frequency pulse trains. 
     FIG. 5C shows a circuit where two inverters, 114 and 116 are coupled. Varying a voltage Vc 5  applied to an input 118 varies resistance across a depletion-type MOSFET 102 and a depletion-type MOSFET 104 which both function as variable current sources An input node 98 and an input node 100 are coupled to outputs nodes of a prior inverter. A voltage V 100  on input node 100 is an inverse of a voltage v 98  on input node 98, i.e., when v 98  is at logic 1, then V 100  is logic 0, andvice-versa. A depletion-type MOSFET 106 and an enhancement-type MOSFET 110 of inverter 114, are coupled as shown to a depletion-type MOSFET 108 and an enhancement-type MOSFET 112 of inverter 116. An output node 94 and an output node 96 are then available to be coupled to a succeeding inverter. 
     The major difference between the circuit in FIG. 5B and the circuit in FIG.5C is the addition of MOSFETs 102 and 104. These MOSFETs were added in inverters 114 and 116, and similar MOSFETs may be added to every inverter in a series of cascaded inverters, to vary the delay time across each inverter by controlling the current through the inverters. This capacity to vary delay time can be used to standardize propagation time between integrated circuits. Propagation time through integrated circuits may varybecause of process variations in the process used to fabricate the integrated circuit. Process variations may include variations in doping density for depletion mode load devices, variations in gate geometries forsmall geometry devices, and environmental variations, such as variations intemperature. 
     FIGS. 5D and 5E illustrate alternative embodiments for coupling inverters. These embodiments allow for compensation for for wide variation in propagation time resulting from process variations, without impingement onother inverter performance parameters. In FIG. 5D each inverter 355 and 356has two depletion-type MOSFETS--MOSFETS 351 and 353 in inverter 355, and MOSFETs 352 and 354 in inverter 356--coupled in series to provide time delay variation through current control. In FIG. 5E each inverter 385 and 386 has one depletion-type MOSFET 381 and an enhancement-type MOSFET--a depletion-type MOSFET 381 and an enhancement-type MOSFET 383 in inverter 385 and a depletion-type MOSFET 382 and an enhancement-type MOSFET 384 in inverter 386--coupled in parallel to provide time delay variance through current control.