Abstract:
A method of interfacing a LC sensor with a control unit is provided. The control unit may include first and second contacts, where the LC sensor is connected between the first and the second contact. A capacitor is connected between the first contact and a ground. To start the oscillation of the LC sensor, the method may include during a first phase, connecting the first contact to a supply voltage and placing the second contact in a high impedance state such that the capacitor is charged through the supply voltage. During a second phase, the first contact may be placed in a high impedance state, and the second contact connected to the ground such that the capacitor transfers charge towards the LC sensor. During a third phase, the first contact and the second contact may be placed in a high impedance state so the LC sensor is able to oscillate.

Description:
TECHNICAL FIELD 
       [0001]    Embodiments of the present disclosure relate to techniques for interfacing an LC sensor. 
       BACKGROUND 
       [0002]    LC sensors are well known in the art. For example, LC sensors may be used as electronic proximity sensors which are able to detect the presence of a conductive target. Some common applications of inductive sensors include, e.g., metal detectors and derived applications, such as rotation sensors. 
         [0003]      FIG. 1  shows the basic behavior of an LC sensor  10 . Specifically, in the example considered, the LC sensor  10  includes an inductor L and a capacitor C, which form a resonant circuit also called tank circuit. The arrangement further includes a power supply  102 , such as a voltage source, and a switch  104 . 
         [0004]    When the switch  102  is in a first position (as shown in  FIG. 1 ), the capacitor C is charged up to the supply voltage. When the capacitor C is fully charged, the switch  102  changes position, placing the capacitor  102  in parallel with the inductor L, and it starts to discharge through the inductor L. This starts an oscillation between the LC resonant circuit  10 . 
         [0005]    From a practical point of view, the LC sensor  10  also includes a resistive component R, which will dissipate energy over time. Accordingly, losses occur which will decay the oscillations, i.e., the oscillation is dampened. 
         [0006]    Such an LC sensor  10  may be used, e.g., to detect metallic objects. This is because the oscillation will be damped quicker in the presence of a metallic object (see, e.g.  FIG. 2   b ) compared to an oscillation without a metallic object (see, e.g.,  FIG. 2   a ). Generally speaking, the sensing component of an LC sensor  10  may be the inductor L, the capacitor C and/or the resistor R. For example, the resistance R primarily influences the damping factor, while the L and C component primarily influence the oscillation frequency. 
         [0007]    Moreover, such a LC sensor  10  may also be created by simply connecting a capacitor C to an inductive sensor L, or an inductor L to a capacitive sensor C. However, the inductor L (with its dissipative losses) usually provides the sensing element. 
         [0008]      FIG. 3   a  shows a possible embodiment for performing the LC sensing of the sensor  10  with a control unit  20 , such as a microcontroller, as described, e.g., in the documents Application Note AN0029 , “Low Energy Sensor Interface - Inductive Sensing ”, Rev. 1.05, 2013-05-09, Energy micro, or Application Report SLAA222A, “ Rotation Detection with the MSP 430  Scan Interface ”, April 2011, Texas Instruments. In the example considered, the control unit  20  has two pins or pads  202  and  204 , and the LC sensor  10  is connected between these pins  202  and  204 . 
         [0009]    The control unit  20  includes a controllable voltage source  206  connected to the pin  202  to impose a fixed voltage V MID  at this pin  202 . For example, a digital-to-analog converter (DAC) is typically used for this purpose. 
         [0010]    During a charge phase, the pin  204  is connected to ground GND. Accordingly, during this phase, the sensor  10  is connected between the voltage V MID  and ground GND, and the capacitor C of the sensor  10  is charged to the voltage V MID . 
         [0011]    Next, the control unit  20  opens the second pin  204 , i.e., the pin  204  is floating. Accordingly, due to the fact that the capacitor C of the sensor  10  has be charged during the previous phase, the LC resonant circuit  10  starts to oscillate as described above. 
         [0012]    Thus, by analyzing the voltage, e.g., voltage V 204  at pin  204 , the oscillation may be characterized. In fact, as shown in  FIG. 3   b , the voltage at the pin  204  corresponds to a damped oscillation having a DC offset corresponding to the voltage V MID , imposed by the voltage source  206 , i.e., the voltage V MID  defines the middle point of the oscillation. Accordingly, the voltage V MID  is usually set to half of the supply voltage of the control unit  20 , e.g., VDD/2, in order to have the maximum range. 
         [0013]    Often, the circuit also includes an additional capacitor C 1  connected between the pin  202  and ground GND to stabilize the voltage signal VMID and to provide the boost of current required to charge the sensor. In order to analyze the signal at the pin  204  (see, e.g.,  FIG. 3   a ), the control unit  20  may include an analog-to-digital converter (ADC)  208  connected to the pin  204  to sample the voltage of the oscillation. Thus, based on the resolution and sampling frequency of the ADC  206 , the whole oscillation may be characterized. 
         [0014]      FIG. 4  shows an alternative approach. More specifically, in the illustrated example, the control unit  20  comprises a comparator  210  which compares the voltage at the pin  204  with a reference signal, such as a reference voltage V Ref . For example, this reference voltage V Ref  may be fixed, e.g., to VDD/2, or set via a digital-to-analog converter  212 . For example,  FIGS. 5   a  and  5   b  respectively show the oscillations with and without a metallic object in the vicinity of the sensor  10 , along with a possible reference voltage V Ref  and the output CMP of the comparator  210 . Generally, the two approaches shown in  FIGS. 3   a  and  4 , i.e., the ADC  208  and comparator  210 , may also be combined in the same control unit  20 . 
         [0015]    Thus, based on the foregoing, contactless motion measurement may be achieved by interfacing LC sensors directly with microcontroller integrated circuits (ICs). Such sensing may be useful, e.g., for metering systems (gas, water, distance, etc.). However, while handling and sampling sensors, microcontrollers (or MCUs) should reduce as much as possible the power consumption to permit the development of battery-powered systems. Moreover, as MCU units are typically general-purpose, there is also the need to reduce as much as possible the silicon area due to the specialized circuits required for the implementation of the above functionality. 
         [0016]    Accordingly, in LC sensor excitation and measurement techniques it is important to reduce consumption and cost, especially for battery powered applications as already mentioned. Thus, a first problem is related to the use of dedicated low power analog components, e.g., for generating the voltage V MID  and the internal reference voltage V Ref , which results in a greater cost. 
         [0017]    A second problem is related to the digital-to-analog converter  210 , that should be both low power and fast enough to follow the dumped oscillation. This leads to a significant power consumption per measurement, and challenging application constraints in battery-powered systems. 
         [0018]    Furthermore, Process-Voltage-Temperature (PVT) variations are another important issue in battery-powered systems, where there are significant voltage changes. Indeed, most of the components described in the foregoing could be affected by the PVT variations, including: sensors (damping factor, frequency, etc.); I/O pads current and resistance (excitation); comparators switching point, etc. 
       SUMMARY 
       [0019]    Based upon the foregoing, there is a need for approaches which overcome one or more of previously outlined drawbacks. 
         [0020]    Such an object is achieved through a method having the features specifically set forth in the claims that follow. A related system is provided, as well as a corresponding related computer program product, loadable in the memory of at least one computer and including software code portions for performing the steps of the method of the invention when the product is run on a computer. As used herein, reference to such a computer program product is intended to be equivalent to reference to a computer-readable medium containing computer-readable instructions for controlling a computer system to coordinate the performance of the method of the invention. Reference to “at least one computer” is intended to highlight the possibility for the present invention to be implemented in a distributed/modular fashion. The claims are an integral part of the disclosure of the invention provided herein. The claims are an integral part of the technical teaching of the invention provided herein. 
         [0021]    As mentioned above, the present description provides approaches for interfacing a LC sensor with a control unit, such as a microcontroller, where the control unit comprises a first and a second contact, such as the pins or pads of a microcontroller. In particular, the LC sensor is connected between two contacts and an additional capacitor is connected between the first contact and a ground. 
         [0022]    In some embodiments, the oscillation of the LC sensor is started by three phases. More specifically, during the first phase, the first contact is connected to a supply voltage and the second contact is placed in a high impedance state, e.g., disconnected, such that the capacitor is charged through the supply voltage provided at the first contact. During the second phase, the first contact is placed in a high impedance state, e.g., disconnected, and the second contact is connected to ground, whereby the capacitor is connected in parallel with the LC sensor and charge is transferred from the capacitor towards the LC sensor. During the third phase, both contacts are placed in a high impedance state, such that the LC sensor is able to oscillate. Accordingly, the oscillation of the LC sensor may be started with a three state driver circuitry, e.g., of a microcontroller. 
         [0023]    In some embodiments, the duration of the second phase, i.e., the charge transfer phase, is varied to regulate the voltage at the capacitor at the beginning of the third phase (i.e., the oscillation phase). This defines the middle point voltage of the oscillation occurring at the second contact. 
         [0024]    In some embodiments, the voltage at the second contact is monitored at least during the third phase to determine some characteristics of the oscillation of the LC sensor. For example, the voltage at the second contact may be compared with at least one reference voltage in order to generate a comparison signal. In this case, the number of pulses in the comparison signal may be counted to characterize the oscillation. Accordingly, the oscillation of the LC sensor may be monitored with an input sensing circuitry, e.g., of a microcontroller. 
         [0025]    In some embodiments, the number of pulse is also used to regulate the middle point voltage of the oscillation occurring at the second contact. For example, this may be done by varying the duration of the second phase, i.e., the charge transfer phase. 
         [0026]    In some embodiments, the charge or discharge behavior of the capacitor may be analyzed via a comparator with hysteresis during a calibration phase. The middle point voltage of the oscillation occurring at the second contact may be regulated by recharging or discharging the capacitor between the second and third phase based on the charge or discharge behavior of the capacitor determined during the calibration phase. Accordingly, the middle point voltage of the oscillation may also be regulated with the three state driver circuitry, e.g., of a microcontroller. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0027]    Embodiments of the present disclosure will now be described with reference to the annexed drawings, which are provided by way of non-limiting example, and in which: 
           [0028]      FIG. 1  is a schematic diagram of an LC sensor in accordance with the prior art; 
           [0029]      FIGS. 2   a  and  2   b  are graphs of voltage oscillation of the LC sensor of  FIG. 1  with and without a metallic object present, respectively; 
           [0030]      FIG. 3   a  is a schematic diagram illustrating another LC sensor arrangement including a controller in accordance with the prior art, and  FIG. 3   b  is a graph of voltage oscillation of the LC sensor of  FIG. 3   a;    
           [0031]      FIG. 4  is a schematic diagram of another LC sensor arrangement including a controller in accordance with the prior art; 
           [0032]      FIGS. 5   a  and  5   b  are graphs of voltage oscillation of the LC sensor arrangement of  FIG. 4  with and without a metallic object present, respectively; 
           [0033]      FIGS. 6 ,  7 ,  9 ,  10  and  12  are schematic diagrams illustrating systems for interfacing an LC sensor in accordance with example embodiments; 
           [0034]      FIGS. 8 ,  14 , and  15  are flow diagrams illustrating methods for interfacing an LC sensor which may be used in the systems of  FIGS. 6 ,  7 ,  9 ,  10  and  12 ; 
           [0035]      FIGS. 11   a - 11   d  and  16  are graphs illustrating exemplary waveforms which may occur in the systems of  FIGS. 6 ,  7 ,  9 ,  10  and  12 ; and 
           [0036]      FIG. 13  is a table providing exemplary results obtained with the systems of  FIGS. 6 ,  7 ,  9 ,  10  and  12 . 
       
    
    
     DETAILED DESCRIPTION 
       [0037]    In the following description, various specific details are given to provide a thorough understanding of embodiments. The embodiments may be practiced without one or several specific details, or with other methods, components, materials, etc. In other instances, well-known structures, materials, or operations are not shown or described in detail to avoid obscuring aspects of the embodiments. 
         [0038]    Reference throughout this specification to “one embodiment” or “an embodiment” means that a particular feature, structure, or characteristic described in connection with the embodiment is included in at least one embodiment. Thus, the appearances of the phrases “in one embodiment” or “in an embodiment” in various places throughout this specification are not necessarily all referring to the same embodiment. Furthermore, the particular features, structures, or characteristics may be combined in any suitable manner in one or more embodiments. The headings provided herein are for convenience only and do not interpret the scope or meaning of the embodiments. 
         [0039]    In the following  FIGS. 6 to 16 , parts, elements or components which have already been described with reference to  FIGS. 1 to 5  are denoted by the same references previously used in such figures. As such, the description of such previously described elements will not be repeated in the following description. 
         [0040]    The embodiments described herein provide approaches that permit an efficient handling of at least one LC sensor  10  by reducing the required dedicated on-chip components and/or by providing reduced power consumption. Some embodiments may also be implemented in a full digital manner with a conventional low cost microcontroller, thus reducing cost. 
         [0041]    Various embodiments may provide an improved resilience against PVT variations (particularly suitable for battery powered systems). In some embodiments, the approach is based on two different techniques, namely Capacitive Dynamic Charge Sharing (CDCS) and Self-Tuning Reference (STR). In some embodiments, such an approach applies a capacitive dynamic charge sharing to remove the V MID  generator and the Vref Generator  206 / 212  shown with respect to  FIGS. 3   a  and  4 , and to use a Self-Tuning Reference technique to permit the use of a fixed internal reference and to improve robustness against PVT variations. 
       Capacitive Dynamic Charge Sharing 
       [0042]    As mentioned above, the Capacitive Dynamic Charge Sharing (CDCS) technique allows for the removal of the V MID  voltage generator module. More particularly, this approach is based on the fact that, in a very short time, the inductance L of the sensor  10  is such that the capacitor C 1  and the capacitor C of the sensor  10  are connected in series. 
         [0043]      FIG. 6  shows the basic architecture of this approach. More specifically, in the illustrated embodiment the LC sensor  10  is connected again (e.g., directly) between the pins  202  and  204  of the control unit  20 , such as a microcontroller. Moreover, a capacitor C 1  is connected (e.g., directly) between the pin  202  and ground GND. As will be described further below, the capacitor C 1  is used in a different manner as compared to the prior art approaches described with respect to  FIGS. 3   b  and  4 . 
         [0044]    In the illustrated embodiment, the control unit  20  does not include a dedicated DAC for generating the voltage V MID , but the control unit  20  merely includes a switch  220  configured to selectively connect the pin  202  to a fixed voltage, such as the supply voltage VDD of the control unit  20  or a voltage signal provided by an internal voltage reference generator, which is often available in conventional microcontrollers. Generally speaking, the supply voltage VDD may be received via a power supply pin of the control unit  20  (not shown). Accordingly, the pin  202  may be either floating or connected to a supply voltage. For example, in some embodiments the operation of the switch  202  may be implemented with a convention three state driver circuitry, e.g., “1” for VDD, “0” for GND and “Z” for a high impedance state, which is often used for output pins of microcontrollers or other digital integrated circuits. 
         [0045]    In the present embodiment, the control unit  20  includes a further switch  222  configured to connect the pin  204  selectively to ground GND. Thus, the operation of the switch  222  may be implemented also with the conventional driver circuitry of an output pin of a microcontroller. 
         [0046]    The switching of the switches  220  and  222  is controlled by a processing unit  230 , such as a digital processing unit programmed via software instructions. For example, this may be the central processing unit (CPU) of a microcontroller or a dedicated digital IP. Accordingly, in some embodiments (see, e.g.,  FIG. 7 ), the above-described driving of the pads  202  and  204  may be implemented with conventional three state driving circuits  240  and  242 , e.g., of a microcontroller  20 . 
         [0047]      FIG. 8  shows a flow chart of the main operations performed by the control unit  20  to start an oscillation of the LC sensor  10 . After a start step  2000 , the control unit  20  connects in a step  2002  the pin  202  to a supply signal, such as the supply voltage VDD of the microcontroller  20 , and the pin  204  is floating. For example, the processing unit  230  may drive the pin  202  with the logic level “1” and the pin  204  with the logic level “Z”. Accordingly, in the step  2002 , only the capacitor C 1  is connected between the supply voltage VDD and ground GND, and the capacitor C 1  is charged. 
         [0048]    Next, the control unit  20  connects the pin  204  to ground GND in a step  2004 , while the pin  202  is floating. For example, the processing unit  230  may drive the pin  202  with the logic level “Z” and the pin  204  with the logic level “0”. Accordingly, in the step  2004  the sensor  10  is connected in parallel with the capacitor C 1 , and the charge on the capacitor C 1  is transferred at least partially to the capacitor C and generally the sensor  10 , i.e., the charge of the capacitor C 1  is shared with the sensor  10 . 
         [0049]    Next, the control unit  20  opens the second pin  204  in a step  2006 , i.e., both pins  202  and  204  are floating. For example, the processing unit  230  may drive both the pin  202  and the pin  204  with the logic level “Z”. Accordingly, due to the fact that the LC sensor  10  has been charged during the step  2006 , the LC resonant circuit  10  starts to oscillate in the step  2008 , as described above. Finally, the procedure terminates at a step  2010 . 
         [0050]    The driving scheme may also include an optional step  2008 , in which the oscillation is stopped. For example, this might be useful if multiple consecutive measurements have to be performed. As shown in  FIG. 8 , this step  2008  may be performed at the end of a measurement (after step  2006 ) or could be performed at the beginning of a new measurement, e.g. before step  2002 . For example, during the step  2008 , both pads  202  and  204  may be connected to ground, e.g., the processing unit  230  may drive both the pin  202  and the pin  204  with the logic level “0”, in order to discharge the capacitors C 1  and C. 
         [0051]    The above description is applicable to a single sensor  10 . However the system may also be extended to multiple sensors, e.g., by using a single pad  202  and a respective sensing pad  204  for each LC sensor. Generally speaking, the amount of charge transferred during the step  2004  depends on the excitation time T excit , in which the switch  222  remains closed while the switch  220  is opened, i.e., the duration of the step  2004 . 
         [0052]    Basically, if the time T excit  is sufficiently short, the inductor L of the sensor may be assumed open and at the end of the step  2004  the total charge originally stored in the capacitor C 1  will be redistributed between the two capacitors C 1  and C, and the voltage at the capacitors C 1  and C will be given by the capacitor divider formula. For example, in case the two capacitors C 1  and C have the same capacitance and assuming instantaneous charge transfer, the voltage on the capacitor C 1  and the capacitor C would reach half of the voltage supply signal VDD. 
         [0053]    However, it will be appreciated that the charge transfer is indeed not “instantaneous”, e.g., due to resistive loads between the capacitor C and C 1 , and the inductor L cannot be assumed always open during the time T excit . That is the capacitor C 1  will also discharge through the inductor L. As a consequence, the final voltages at the capacitor C 1  and the capacitor C depend on the time T excit , i.e., the voltages reached by the capacitor C 1  and the sensor capacitor C (at the end of the step  2004  and the beginning of step  2006 ) depend on the excitation time T excit . 
         [0054]    Accordingly, the Capacitive Dynamic Charge Sharing (CDCS) technique shown in  FIGS. 6 and 7  is based substantially on a capacitive divider principle (exploiting existing components) applied during a transitory period. Specifically, in the embodiment considered, the capacitor C 1  is pre-charged to VDD, and the charge is transferred partially to the sensor  10  as a function of the duration T excit  of the step  2004 , i.e., while pin  202  is floating and pin  204  is connected to ground. However, as described above, the voltage at the capacitor C 1  during the step  2006 , in which the LC sensor is oscillating, constitutes the middle point voltage V MID  of the oscillation. Accordingly, by controlling the duration T excit , it is possible to regulate the voltage V MID , i.e., the voltage at the capacitor C 1  at the end of the step  2004  or the beginning of step  2006 . 
       Self-Tuning Reference 
       [0055]    The Self-Tuning Reference (STR) technique, when used in conjunction with the previously described Capacitive Dynamic Charge Sharing (CDCS) technique, permits the use of a simple comparator with fixed (e.g., internal) reference value V Ref  to analyze the oscillation during the step  2006 . Accordingly, no digital-to-analog converter (e.g., block  208  in  FIG. 3   a ) or controllable voltage reference (e.g., block  212  in  FIG. 4 ) is required. 
         [0056]    For example, as shown in  FIG. 9 , a comparator  250  may be connected to the pin  204  and compare the voltage at the pin  204  with a fixed reference value V Ref . The result of the comparison CMP may then be made available to the processing unit  230 , e.g., the digital processing core of a microcontroller, which may be configured for analyzing the sequence of pulses in the signal CMP. 
         [0057]    For example, in some embodiments, a comparator with hysteresis, such as a Schmitt Trigger, with fixed thresholds may be used to analyze the oscillation. For example, such Schmitt Triggers with fixed thresholds are often used in the sensing circuitry of the input pads of microcontrollers or other digital integrated circuits. Accordingly, no additional components may be required and the conventional sensing circuitry of an input pin of microcontroller may be used. 
         [0058]    By way of example, as shown in  FIG. 10 , the conventional sensing circuitry  260  of an input pad, e.g., of a microcontroller, may be used to implement the comparator  250 . Accordingly, the result of the comparison may be directly available to the processing core  230  by merely “reading” the value associated with the input pad  204 . 
         [0059]    In the prior-art approach described with respect to  FIG. 4 , the possibility of tuning the internal reference voltage V Ref  via the source  212  usually permits setting a reference value V Ref  which ensures that enough digital pulses are generated at the output CMP of the comparator, but not too many pulses to avoid a waste of time and power (see also  FIGS. 5   a  and  5   b ). Conversely, in some embodiments, the above-mentioned Capacitive Dynamic Charge Sharing technique is used to selectively vary the middle point voltage V MID  of the oscillation instead of the threshold voltage of the comparator  250 . Accordingly, the role of V MID  and V Ref  are swapped, i.e., by moving the Voltage V MID , the number of digital pulses may be varied in a substantially similar way as moving the voltage V Ref . 
         [0060]    By way of example,  FIG. 11   a  shows a typical oscillation of an LC sensor with a middle point V MID  (which usually corresponds to 0.5 VDD) and the reference voltage V Ref , which in the example is set to V MID . Conversely,  FIG. 11   b  shows an example in which the middle point voltage V MID  has been raised to change the number of digital pulses instead of moving the voltage V Ref . 
         [0061]    Similarly,  FIG. 11   c  shows the waveform of  FIG. 11   a , in which a Schmitt Trigger has been used, e.g., with a lower threshold TL of 0.4 VDD and a upper threshold TH of 0.6 VDD. Finally,  FIG. 11   d  shows the waveform of  FIG. 11   b  with raised middle point voltage V MID , and where the Schmitt Trigger of  FIG. 11   c  has been used. 
         [0062]    As shown in the above  FIGS. 11   a  to  11   d , the number of pulses at the output of the comparator  210  varies for the same waveform as a function of the middle point voltage V MID . However, as mentioned above, the middle point voltage V MID  varies as a function of the excitation time T excit  during the charge transfer phase  2004 . Thus, by controlling the time T excit , the comparison result may be tuned. 
         [0063]      FIG. 12  shows in this context an embodiment of an integrated circuit  20 , such as a microcontroller, which may be used to perform the above-noted operations. More specifically, pad  204  is an input and output pad with the associated three state output drive circuitry  242  and input sensing circuitry  260 , such as a Schmitt Trigger. Pad  202  is at least an output pad with the associated three state output drive circuitry  240 . 
         [0064]    Accordingly, by driving the pads  202  and  204  via the driver circuitry  240  and  242  as described above, in particular with respect to  FIG. 8 , the oscillation of the LC sensor  10  may be stimulated and the middle point voltage V MID  may be set. More specifically, the driving of the pads  202  and  204  may be performed via the digital processing core  230 . 
         [0065]    Once the oscillation has been started, the output from the sensing circuitry  260  is fed to the processing core  230  for further analysis to determine characteristics of the oscillation. For example, as shown with respect to  FIGS. 5   a  and  5   b , the output CMP is indicative for the damping factor of the oscillation, which in turn is indicative for the presence of a metallic object near the sensor  10 . Generally speaking, the digital processing unit  230  may be a dedicated hardware module, a general-purpose processor programmed via software instructions or a combination of both. 
         [0066]    Thus, counting of the pulses in the signal CMP may also be performed via the digital processing core. However, the oscillation may have a high frequency, in which case counting via software instructions may not be feasible. Accordingly, such this case the control unit  20  may include a hardware-implemented counter  270 , which already is often included in conventional microcontrollers, and the output of the sensing circuitry  260  may be fed to the counter  270 . Thus, the counter  270  may count the number of pulses in the signal CMP independently from the processing unit  230  and the processing unit  230  may only read the final result, i.e., the signal at the output of the counter  270 , and eventually reset the counter  270  when a new measurement is started. 
         [0067]    Moreover, the counter  270  may also be extended to provide a dedicate measurement and processing unit which directly elaborates the signal CMP to extract the information required. For example, the measurement and processing unit  270  may directly detect the sensor&#39;s state, such as over metal, over plastic, etc. 
         [0068]    The module  270  may also generate at least on programmable interrupt on specific conditions. For example, such a measurement and processing unit may also be connected to a plurality of sensing pads  204  to elaborate the signal from a plurality of sensors, e.g., to perform a speed or rotation measurement. 
         [0069]    As shown with respect to  FIGS. 11   a  to  11   d , the number of pulses at the output of the comparator  210  varies for the same waveform as a function of the middle point voltage V MID . The middle point voltage V MID  in turn varies as a function of the excitation time T excit  during the charge transfer phase  2004 . 
         [0070]    In some embodiments, the Self-Tuning Reference (STR) technique looks directly at the number of digital pulses generated at the output of the comparator, e.g. the Schmitt Trigger  260  of  FIG. 10  to automatically tune the excitation time T excit , to be used in the CDCS technique described in the foregoing. In this way, a desired number of digital pulses may be achieved, which usually corresponds to a given reference condition (e.g., with metal). For example, the reference condition usually corresponds to the situation with the greatest damping factor, which corresponds to the oscillation with the lowest expectable number of pulses in the output CMP of the comparator  250 / 260 . By way of example, in some embodiments a closed-loop regulation is used to set the time T excit  to ensure that the number of pulse for a given reference condition, e.g., the condition with the greatest damping factor, corresponds to the target number of pulses K. In this case, when measuring the reference condition the number of pulses at the output of the comparator will include K counts, and the number of pulses will increase in condition with a lower damping factor. 
         [0071]    For example, considering an exemplary case where the resistance R in the sensor  10  (which primarily models the damping behavior) may be between 3 and 45 Ohms, and the minimum number of count K should be 4, the calibration would be performed for the condition with R=45 Ohm. By way of example, for a typical LC sensor, the final results may then be: 
         [0072]    4 pulses for R=45 Ohm; 
         [0073]    5 pulses for R=37 Ohm; and 
         [0074]    9 pulses for R=3 Ohm. 
         [0075]    Moreover, the described calibration mechanism renders the system robust against variations of parameters which influence the oscillation. For example,  FIG. 13  shows a table including the number of pulses in the signal CMP for different supply voltages VDDε{3.3V, 2.V, 2.5V, 2.1V}, temperatures Tε{−30° C., 25° C., 125° C.}, and resistances Rε{3 Ohm, 37 Ohm, 45 Ohm}. As shown in  FIG. 13 , this approach is very robust against voltage variations, while the resolution may be affected by low temperatures. 
         [0076]    In some embodiments, instead of performing the calibration only once, the Self-Tuning Reference technique may be run continuously and regulate the voltage V MID , ensuring that the number of pulses in the signal CMP for a measurement is never smaller than K. For example, this may be useful for rotation sensors where a disc with a metal profile is rotated in front of at least one LC sensor  10 , because in this case it may be difficult to establish a priori the correct reference condition. Thus, generally speaking, the Self-Tuning Reference technique may be performed by the digital processing unit  230  or also directly by the measurement and processing unit  270 . 
         [0077]    The STR technique may also be used to identify the direction to take when modifying the time T excit  and/or cope with deadlocks, which may occur when the time T excit  is out of the valid range. For example, in some embodiments, the following parameters may be used:
       NP—number of pulses for the current measurement cycle;   PNP—number of pulses for the previous measurement cycle;   DIR—direction;   PDIR—previous direction;   K—target minimum number of pulses for a measurement;   T excit —excitation time, e.g. in clock periods during which the capacitor C 1  transfers charge to the sensor  10 ; and   TO—timeout, e.g., in measurement cycles.       
 
         [0085]    In some embodiments, when the number of measured pulses NP is less than the target K and less than pulses in the previous cycle PNP, a direction change may be forced, because it may be assumed that the time T excit  should be corrected in the opposite direction. In some embodiments, a counter C is used to check whether the timeout condition occurs. For example, such a counter C may be incremented each time the number of measure pulses NP is less than K but equal to the previous one NPN. Accordingly, if this condition is true for TO measurement cycles, the parameter T excit  is out of range, because there is no more sensitivity to a variation of T excit . For example, in this case, the time T excit  may be reset to its original value and the direction is changed. 
         [0086]    By way of example,  FIG. 14  shows a flow chart of a method which may be used to automatically determine the time T excit . After a start step  3000 , the procedure starts and the parameters are initialized in a step  3002 . For example, in this step  3002  the counter C may be reset (e.g., set to zero), the parameter PNP is set to zero, and the time T excit  is set to a initial default value (e.g., zero). 
         [0087]    The procedure continues at a step  3004  where a measurement is performed. If the calibration procedure is always switched on, the procedure may also merely monitor whether a measurement has been performed. 
         [0088]    In a verification step  3006 , the procedure verifies whether the measured number of pulses NP is less than the target value K. If the measured number of pulses NP is equal or greater than the target K (output “N” of the conditional step  3006 ), no correction is required and the procedure continues at a step  3008  where the timeout counter C is reset (e.g., set to zero), and the procedure returns to step  3004 . 
         [0089]    On the contrary, where the measured number of pulses NP is less than the target value K (output “Y” of the verification step  3006 ), some correction may be required and the procedure continues at a step  3010 . Specifically, in the verification step  3010 , the procedure verifies whether the measured number of pulses NP is less than the previous number of pulses PNP. 
         [0090]    When the measured number of pulses NP is less than the previous number of pulses PNP (output “Y” of the verification step  3010 ), the direction DIR for the correction of the time T excit  is inverted at a step  3012 . For example, if the previous direction PDIR indicates that the time T excit  should be decreased, the new direction DIR indicates now that the time T excit  should be incremented. On the contrary, if the previous direction PDIR indicates that the time T excit  should be incremented, the new direction DIR indicates now that the time T excit  should be decremented. 
         [0091]    Moreover, in this case the counter C is reset at a step  3014 , and the time T excit  is updated at a step  3016 , e.g., by decrementing or incrementing the value of T excit  based on the updated parameter DIR. For example, in an example embodiment the parameter T excit  is varied merely by one clock cycle, i.e., T excit =T excit ±1. However, the variation may depend on the velocity of the control unit, e.g., the frequency of the clock signal. 
         [0092]    Finally, the parameters of the previous cycle are update at a step  3018 , e.g., by assigning the value of the direction DIR to the previous direction PDIR and the value of the number of pulses NP to the previous number of pulses PNP. On the contrary, if the measured number of pulses NP is equal or greater than the previous number of pulses PNP (output “N” of the verification step  3010 ), the direction DIR for the correction of the time T excit  is usually correct. 
         [0093]    However, in this case it may be verified whether a timeout condition is reached. For example, in the embodiment considered, the procedure verifies whether the measured number of pulses NP is equal to the previous number of pulses PNP in a step  3020 . 
         [0094]    More specifically, if the number of measured pulses NP is not equal to the previous number of pulses PNP (output “N” of the verification step  3020 ) and taking into account that is has previously been verified that the measured number of pulses NP is not smaller than the previous number of pulses PNP (see step  3010 ), the measured number of pulses NP is greater than the previous number of pulses PNP. Accordingly, in this case the correction is going in the correct direction and the timeout counter C may be reset and the time T excit  may be updated, i.e., incremented or decremented based on the current direction DIR. For example, in the present embodiment, the procedure simply proceeds at the step  3014  for this reason. 
         [0095]    Conversely, in case the where number of measured pulses NP is equal to the previous number of pulses PNP (output “Y” of the verification step  3020 ), a timeout condition may be present. This is because the last variation of the time T excit  did not influence the measured number of pulses. 
         [0096]    Accordingly, in some embodiments, the procedure continues to increment or decrement the time T excit  until a variation of the number of pulse occurs or a timeout is reached. For example, in the present embodiment, the procedure continues for this reason at a verification step  3022 , in which the procedure verifies whether the counter C has reached the timeout value TO. 
         [0097]    When the counter C has not reached the timeout value TO (output “N” of the verification step  3022 ), a single variation of the time T excit  might have been insufficient, and the counter C is incremented in a step  3024 . Moreover, in this case the procedure continues to vary the time T excit  in the current direction, i.e., incremented or decremented T excit  based on the current direction DIR. For example, in the present embodiment, the procedure proceeds at the step  3016  for this reason. 
         [0098]    Conversely, if the counter C has reached the timeout value TO (output “Y” of the verification step  3022 ), a timeout condition occurred, i.e., variations of the time T excit  do not influence anymore the number of pulses. In this case, a possible approach may be to see if variations in the opposite direction are suitable to reach the required number of pulses K. For example, in an example embodiment, the direction is inverted and the time T excit  is set to the previous value before a timeout condition was reached. 
         [0099]    In the present embodiment, the direction DIR for the correction of the time T excit  may be inverted at a step  3026 . For example, if the previous direction PDIR indicates that the time T excit  should be decreased, the new direction DIR indicates now that the time T excit  should be incremented. On the contrary, if the previous direction PDIR indicates that the time T excit  should be incremented, the new direction DIR indicates now that the time T excit  should be decremented. 
         [0100]    Moreover, in this case the counter C is reset at a step  3028 , and the time T excit  is set to the previous value T excit  at a step  3030 . For example, if the new direction DIR indicates that the time T excit  should be incremented, the timeout value TO may be added to the time T excit , i.e. T excit =T excit +TO, thus turning back to the value of T excit  prior to the timeout loop. On the contrary, if the new direction DIR indicates that the time T excit  should be decremented, the timeout value TO may be subtracted from the time T excit , i.e., T excit =T excit −TO. 
         [0101]    Finally, the procedure may continue in this case at step  3018  to update the parameters of the previous cycle. For example, the convergence of the above described procedure has been verified with a conventional microcontroller for K=4 and TO=4. 
         [0102]    In addition to the above-described methods for setting the minimum number of pulses K, a different approach may also be used to set the time T excit . More specifically, in some embodiments, the voltage V MID  is determined via a Schmitt Trigger connected to pad  202 , e.g., a respective input circuitry  262  of the pad  202  (see, e.g.,  FIG. 12 ) similar to the one described for the pad  204 . 
         [0103]    In an example embodiment, by driving the pads  202  and  204  and by monitoring the voltage at the pad  202  via a Schmitt trigger, it is possible to regulate the voltage V MID . More specifically,  FIG. 15  shows a calibration procedure and  FIG. 16  shows a respective waveform of the voltage at pad  202 , and thus the voltage V MID  at the capacitor C 1 , for a given period of time t. 
         [0104]    After a start step  4000 , the control unit  20  sets in a step  4002  the pad  202  to the voltage VDD and the pad  204  to a high impedance state. For example, the processing unit  230  may drive the pin  202  with the logic level “1” and the pin  204  with the logic level “Z”. 
         [0105]    Accordingly, this condition corresponds to step  2002  described above with respect to  FIG. 8 . That is, only the capacitor C 1  is connected between the supply voltage VDD and ground GND and the capacitor C 1  is charged. 
         [0106]    Once the voltage V 202  at the pad  202  is stable (e.g., after a given period of time), the control unit  20  connects in a step  4004  (at time t 1 ) the pad  204  to ground GND and sets the pad  202  to a high impedance state. For example, the processing unit  230  may drive the pin  202  with the logic level “Z” and the pin  204  with the logic level “0”. Accordingly, this condition corresponds to step  2004  described with respect to  FIG. 8 , in which the sensor  10  is connected in parallel with the capacitor C 1  and the charge on the capacitor C 1  is transferred at least partially to the sensor  10 . Accordingly, in this stage the voltage at pad  202  decreases as shown in  FIG. 16 . 
         [0107]    In the present embodiment, the processing unit  230  monitors the logic level CMP 202  at the output of the Schmitt Trigger  262  associated with the pad  202 . In fact, while the voltage V 202  remains above the lower threshold TL of the Schmitt Trigger, the signal CMP 202  will be high, i.e., the logic level “1”. 
         [0108]    At the moment t 2  when the signal CMP 202  goes low, i.e., the logic level “0”, the voltage V 202  has reached the lower threshold TL. Immediately after having detected that the signal CMP 202  has gone low, i.e., at the instant t 2 , the control unit  20  sets at step  4006  the pad  202  to the voltage VDD and the pad  204  is connected to Z. 
         [0109]    Accordingly, at time t 1  the capacitor C 1  stored the following charge: 
         [0000]        Q   t1   =C 1· Vdd,  
 
         [0000]    while the capacitor C 1  stored only the following charge at the time t 2 : 
         [0000]        Q   t2   =C 1· TL,  
 
         [0000]    i.e., the following charge has been transferred to the LC sensor  10 : 
         [0000]        Q   LC   =Q   t1   −Q   t2 , 
         [0110]    Accordingly, at this moment the oscillation of the LC sensor  10  has been started and the pin  202  could also be disconnected or placed in a high impedance state. Conversely, in the embodiment considered, at this stage the capacitor C 1 , (i.e., pin  202 ) is connected again to the supply voltage VDD to recharge the capacitor C 1 , thus increasing the middle point voltage V MID . By way of example, the processing unit  230  may drive the pin  202  with the logic level “1” and the pin  204  with the logic level “Z”. 
         [0111]    At the moment t 3  when the signal CMP 202  goes to high, (i.e., the logic level “1”), the voltage V MID /V 202  has reached the upper threshold TH. Thus, the time between t 2  and t 3  is indicative for the time required to charge the capacitor C 1  from TL to TH. 
         [0112]    Accordingly, the control unit  20  may detect during the calibration phase in a step  4008  the time elapsed between the instants t 2  and t 3  and perform during the normal operation a recharging with a recharge time T recharge  determined as a function of the time elapsed, thus regulating the middle point voltage V MID  to be used during the normal operation. For example, the maximum number of pulses in the signal CMP may be expected by setting the recharge time to: 
         [0000]        T   recharge =( t 3− t 2)/2,
 
         [0000]    because in this case, the middle point voltage V MID  should correspond more or less to: 
         [0000]        V   MID =( TH−TL )/2. 
         [0113]    For example, in some embodiments, the method shown in  FIG. 8  is modified for this purpose, e.g., by adding an additional step between the step  2004  and the step  2006 . Specifically, once the comparison signal CMP 202  indicates that the voltage V 202  at the first contact  202  is below the lower threshold TL, the first contact  202  is connected again to the supply voltage VDD such that said capacitor C 1  is recharged through the supply voltage VDD. More specifically, the recharge duration T recharge  of the capacitor C 1  is determined as a function of the duration of the above duration t 3 −t 2  of the calibration phase  4006 , thereby defining the middle point voltage V MID . Finally, the procedure terminates at a stop step  4010 . 
         [0114]    In some embodiments, instead of monitoring the recharge time between the thresholds TL and TH (i.e., t 2  and t 3 ), the procedure may monitor the discharge time between the thresholds TH and TL. For example, in an example embodiment, the procedure may again discharge the capacitor C 1  after the step  4006 , e.g., by using the driving describe with respect to step  4004 . That is, once the voltage V 202  has reached the threshold TH and the logic level goes to high, the pad  204  is connected to ground GND and the pad  202  is set to a high impedance state. Thus, by monitoring the time when the lower threshold TL is reached, i.e., when the logic level of CMP 202  goes to low, it is possible to determine the discharge behavior and set the discharge time T discharge  accordingly. 
         [0115]    Generally speaking, this calibration procedure may also be performed periodically. Moreover, in some embodiments, the previously-described closed loop calibration methods (e.g., the method for setting the time T excit  described with respect to  FIG. 14 ), may also be used to regulate the times T recharge  or T discharge . 
         [0116]    Accordingly, as described above, the Self-tuning Reference technique takes advantage of moving the external reference voltage V MID  to avoid a variable internal reference signal. While the embodiments have been described in combination with the CDCS technique, generally speaking, this approach may be applied also to prior art approaches, in which the middle point voltage V MID  is imposed via a voltage signal (see, e.g.,  FIG. 3   a ). Therefore, the Self-Tuning Reference (STR) technique automatically tunes the time T excit  or directly the middle point voltage V MID  to meet a target number of pulses regardless of the working parameters (and in general PVT variations). 
         [0117]    The details of construction and the embodiments may vary with respect to what has been described and illustrated herein purely by way of example, without departing from the scope of the present disclosure, as defined by the ensuing claims.