Abstract:
A control device for a switching power converter having an inductor element, a switch coupled to the inductor element, a storage element coupled to an output on which an output voltage is provided, and a diode element coupled to the storage element. The control device generates a command signal to control the switch and determine storage of energy in the inductor element in a first interval, and transfer of energy onto the storage element through the diode element in a second interval. A voltage shifter module generates a feedback voltage shifted relative to the output voltage. An amplification module has a first input receiving the feedback voltage, a second input receiving the reference voltage, and an output that supplies, as a function of the difference between the feedback and reference voltages, a control signal. A control unit receives the control signal and generates the command signal to control the switch.

Description:
BACKGROUND 
     Technical Field 
       [0001]    The present disclosure relates to a device and to a method for closed-loop control of a power converter, in particular a DC/DC voltage converter. 
       Description of the Related Art 
       [0002]    As it is known, power converters, in particular voltage converters, are widely used in electronic systems for performing voltage step-up or voltage step-down operations. 
         [0003]    In particular, voltage converters, typically with a voltage step-up function, are used in microelectromechanical systems (MEMS), for raising a low-voltage (LV) input voltage coming from a battery (or from a similar supply source) into a high-voltage (HV) output voltage, used, for example, for driving piezoelectric electrodes or elements of related microelectromechanical structures and causing actuation of inertial masses thereof, for example, by the electrostatic or piezoelectric effect. 
         [0004]    As shown in  FIG. 1 , a power converter  1 A, which may also be referred to as a switching power converter or simply a switching converter herein, is of a voltage-booster (or step-up) type, in this example of a flyback type, generally including a transformer element  2 , having: a primary winding  2   a , with a first winding terminal connected to an input IN of the power converter  1 A, which receives a low-voltage input voltage V IN  (for example, of the order of a few volts, for example, between 3 and 5 V), and a second winding terminal connected to a first internal node N 1 ; and a secondary winding  2   b , having a respective first winding terminal coupled via a diode element  5  to an output OUT of the power converter  1 A, which supplies a high-voltage output voltage V OUT  (for example, of the order of a few tens of volts, e.g., between 40 and 200 V), and a respective second terminal connected to a reference terminal or ground terminal (GND). 
         [0005]    In particular, the aforesaid diode element  5  has its anode terminal connected to the first winding terminal of the secondary winding  2   b  of the transformer element  2 , and its cathode terminal connected to the output OUT of the power converter  1 A. 
         [0006]    The power converter  1 A further comprises: a switch element  4 , in particular made by a transistor, in the example a MOSFET, having a first current-conduction terminal connected to the second winding terminal of the primary winding  2   a  of the transformer element  2 , a second current-conduction terminal connected to the reference terminal GND, and a control terminal that receives a command signal S GD ; and a capacitor element  6 , connected between the output terminal OUT and the reference terminal GND. 
         [0007]    The power converter  1 A further comprises a control device  8 A, designed to generate the command signal S GD , for controlling switching of the switch element  4  and thus regulating the value of the output voltage V out . 
         [0008]    The control device  8 A receives at its input a feedback voltage V FB , indicative of the value of the output voltage V out , in particular generated on a feedback node FB by a resistive divider  9  coupled to the output OUT and constituted by: a first voltage-divider resistor  9   a , connected between the feedback node FB and the reference terminal gnd, and a second voltage-divider resistor  9   b , connected between the feedback node FB and the output OUT. In the embodiment illustrated, the second voltage-divider resistor  9   b  is formed by a plurality of resistor elements  9   b ′, electrically connected in series between the feedback node FB and the output OUT. 
         [0009]    The feedback voltage V FB  is consequently a signal with values in the low-voltage interval, generated by division of the output voltage V out , i.e., a signal with high-voltage values. 
         [0010]    The control device  8 A, in the embodiment illustrated, is configured so as to implement a hysteresis control, i.e., so as to control the value of the output voltage V out  between a lower threshold value V TH   −  and an upper threshold value V TH   + . 
         [0011]    In detail, the control device  8 A comprises a comparator  10 , having a first comparison input connected to the feedback node FB and receiving the feedback voltage V FB , a second comparison input receiving the lower threshold value V TH   −  and upper threshold value V TH   + , and a comparison output, which supplies a comparison signal S comp , as a function of the results of the comparison with hysteresis between the feedback voltage V FB  and the lower threshold value V TH   −  or upper threshold value V TH   + , as illustrated in the diagrams of  FIG. 2 . 
         [0012]    The control device  8 A further comprises a logic unit  11 , coupled to the output of the comparator  10 , which receives the comparison signal S comp  and generates the command signal S GD  on the basis of the same comparison signal S comp . 
         [0013]    The present Applicant has realized that this control solution, albeit functional, at least in some operating conditions has some drawbacks. 
         [0014]    In particular, the errors of the comparator  10  and the errors possibly present on the voltage references, i.e., on the lower threshold value V TH   −  and upper threshold value V TH   + , are amplified by the voltage-division gain introduced by the voltage divider  9 , which generates the value of the output voltage V out ; in other words, these errors are amplified in the output signal V out . Furthermore, the same low-voltage hysteresis is multiplied by the voltage-division gain. 
         [0015]    The electrical characteristics of the comparator  10  consequently substantially affect proper regulation of the value of the output voltage V out  (in terms of offset value, regulation rate, and so forth). 
         [0016]      FIG. 3  illustrates a further known embodiment of a control device, designated by  8 B in  FIG. 3 , of a power converter  1 B (by way of example, once again of a flyback type, made in a manner similar to what has been discussed with reference to  FIG. 1 ). 
         [0017]    The control device  8 B in this case is configured to implement a closed-loop continuous-time control of the value of the output voltage V out  and thus comprises: an operational amplifier  12 , in particular an operational transconductance amplifier (OTA), having a first input terminal that receives a reference voltage V ref , with an appropriate value (for example, generated by a band-gap generator starting from a supply voltage of the power converter  1 B), a second input terminal connected to the feedback node FB by a first impedance element Z 1 , and an output, which is connected to the second input terminal by a second impedance element Z 2  and supplies a control voltage V c , as a function of the error between the reference voltage V ref  and the feedback voltage V FB , as illustrated in the diagrams of  FIG. 4 . The control voltage V c  is useful for controlling the value of the output voltage V out . 
         [0018]    The control device  8 B further comprises a PWM control unit  13 , which receives the control voltage V c  and generates in an appropriate way the command signal S GD  for the switch element  4  in such a way as to minimize the aforesaid error between the reference voltage V ref  and the feedback voltage V FB . 
         [0019]    In particular, in a known way, the PWM control unit  13  may implement a voltage control mode or a current control mode, on the basis of the same control voltage V c , for generating the aforesaid command signal S GD , and in particular for defining the duration of the ON interval (T ON ) and OFF interval (T OFF ) of the switch element  4  during a corresponding switching period. 
         [0020]    For instance,  FIG. 5  shows a power converter  1 C including a possible embodiment of a known type of the PWM control unit  13  of  FIG. 3  in a control device  8 C in the case where the PWM control unit implements a voltage control mode. 
         [0021]    In this case, the PWM control unit  13  comprises a comparator module  15 , having a first comparison terminal coupled to the output of the operational amplifier  12  and receiving the control voltage V C , a second comparison terminal receiving a ramp voltage V R , and an output generating a comparison signal S c . 
         [0022]    In particular, the ramp voltage V R  is generated by charging of a capacitor  16  with an appropriate charging current I c , having a constant value, generated by a current generator  17 . A switch  18 , controlled by a control signal SW, is connected in parallel to the capacitor element  16  in order to determine start of charging and subsequent discharging thereof (and thus the alternately increasing and decreasing pattern of the ramp voltage V R ). 
         [0023]    The PWM control unit  13  further comprises a S/R flip-flop  19 , having: a reset input R, which is connected to the output of the comparator module  15  and receives the comparison signal S c ; a set input S connected to a timing module  20 ; a negated output  Q , which defines the aforesaid control signal SW of switch  18 ; and an output Q, which supplies, via a driver unit  21 , the command signal S GD  for controlling switching of the switch element  4  of power converter  1 C. 
         [0024]    The timing module  20  comprises a detection unit  23 , which, in a known manner, here not described in detail, generates a detection signal ZCD indicative of the occurrence of a zero-current condition in the primary winding  2   a  of the transformer element  2  of the power converter  1 . 
         [0025]    The timing module  20  further comprises: a delay unit  22 , connected to the output of the detection unit  23  and configured to apply an appropriate time delay to the detection signal ZCD; and a OR logic gate  24 , having a first input connected to the output of the delay unit  22 , a second input connected to a starter unit  25 , and an output that supplies the set signal S to the S/R flip-flop  19 . 
         [0026]    In use, assuming that the switch element  4  is initially closed, the ramp voltage V R  grows until it reaches the value of the control voltage V C , thus switching the output of the comparator module  15 , which resets the S/R flip-flop  19  and causes opening of switch element  4 . 
         [0027]    Consequently, the output of the comparator module  15  determines the duration of the ON interval T ON  of the switch element  4 . 
         [0028]    After opening of the switch element  4 , the energy previously stored in the transformer element  2  is transferred onto the capacitor element  6  and onto the load, up to complete demagnetization of the primary winding  2   a.    
         [0029]    The detection unit  23  of the timing module  20  generates a pulse following upon detection of this demagnetization. This pulse in the detection signal ZCD, appropriately delayed by the delay unit  22 , then sets the S/R flip-flop  19  and closes the switch element  4  again, thus determining a new operating cycle of the power converter  1 . 
         [0030]    Accordingly, the control solution envisages that starting of each new switching cycle occurs after the current in the primary winding  2   a  has reached a zero level, following upon an appropriate delay. It should be noted that the presence of the OR logic gate  24  in the timing module  20  and of the aforesaid starter unit  25 , enables start of a new operating cycle, for example, during turning-on (or during return from a de-activation condition). 
         [0031]    In an equally known manner, not illustrated in detail herein in order not to burden the description, in the case of a current control mode, the PWM control unit  13  is configured to generate a control current starting from the control voltage V C  and comparing this control current with a detection current indicative of the current flowing in the primary winding  2   a  (or in the secondary winding  2   b ) of the transformer element  2 . In this case, the command signal S GD  is generated as a function of the comparison between the control current and this detection current. 
         [0032]    The present Applicant has realized that also the further control solution described with reference to  FIG. 3 , at least in some operating conditions, has some drawbacks. 
         [0033]    In the first place, also in this solution, errors of the operational amplifier  12  and errors on the reference voltage V ref  are amplified by the voltage-division gain defined by the voltage divider  9 . 
         [0034]    The accuracy of the regulation of the output voltage V out  markedly depends upon the electrical characteristics of the error amplifier  12  (DC gain, amplifier offsets, and so forth). 
         [0035]    Furthermore, the frequency compensation of the closed control loop in general requires a compensation network. Consequently, the impedance elements Z 1 , Z 2  define in general a complex compensation network, frequently made with components external to the integrated circuit. 
       BRIEF SUMMARY 
       [0036]    Embodiments of the present disclosure solve at least some of the problems highlighted previously, and in particular provide a closed-loop control solution of an improved type for a switching power converter. 
         [0037]    According to embodiments of the present disclosure, a control device of a power converter, and a corresponding control method, are consequently provided. In one embodiment, a control device for controlling a converter of the switching type includes a voltage shifter module configured to be coupled to an output of the converter. The voltage shifter module receives a shifting reference voltage and generates on a feedback node a feedback voltage having a value based on an output voltage on the output and the shifting reference voltage. An amplification module has a first input coupled to the voltage shifter module to receive the feedback voltage and a second input coupled to receive a reference voltage. The amplification module generates on an output a control signal as a function of a difference between the feedback voltage and the reference voltage. A control unit is coupled to the amplification module to receive said control signal and generates a command signal to be applied to control switching of a switch element of the converter as a function of the control signal. 
     
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
         [0038]    For a better understanding of the present disclosure, preferred embodiments thereof are now described, purely by way of non-limiting example and with reference to the attached drawings, wherein: 
           [0039]      FIG. 1  shows a general circuit diagram of a power converter and of a corresponding control device of a known type; 
           [0040]      FIG. 2  shows plots of electrical quantities in the control device of  FIG. 1 ; 
           [0041]      FIG. 3  shows a general circuit diagram of the power converter and of a further control device of a known type; 
           [0042]      FIG. 4  shows plots of electrical quantities in the control device of  FIG. 3 ; 
           [0043]      FIG. 5  shows a more detailed circuit diagram of the control device of  FIG. 3 , which implements voltage control; 
           [0044]      FIG. 6  shows a circuit block diagram of a control device of a switching power converter, according to one embodiment of the present solution; 
           [0045]      FIG. 7  shows a more detailed circuit solution of a voltage-shifter module in the control device of  FIG. 6 ; 
           [0046]      FIG. 8  shows a more detailed circuit solution of an amplifier module in the control device of  FIG. 6 ; 
           [0047]      FIG. 9  shows a more detailed circuit solution of the amplifier module of  FIG. 6  and of an associated voltage control loop; and 
           [0048]      FIG. 10  is a block diagram of a microelectronic system in which the power converter and the corresponding control device are used according to a further aspect of the present solution. 
       
    
    
     DETAILED DESCRIPTION 
       [0049]    With reference to  FIG. 6 , an embodiment is now described of a control device, designated by  30 , of a power converter, which in this case is a voltage converter designated by  1 D and, purely by way of example, made as described with reference to  FIG. 1  and thus being of a flyback booster type. The power converter  1 D is consequently not described again in detail. 
         [0050]    The control device  30  of the power converter  1 D comprises a voltage-shifter module  32 , which is connected to the output OUT of the power converter  1 D, from which it receives the output voltage V out , with a high voltage, and is designed to supply on the feedback node FB the feedback voltage V FB , with a low voltage, according to the expression: 
         [0000]    
       
      
       V 
       FB 
       =V 
       out 
       −V 
       t  
      
     
         [0000]    where V t  is an appropriate shifting value for shifting the output voltage V out  into a low-voltage range. 
         [0051]    In particular, in this embodiment, the voltage-shifter module  32  comprises: a shifting resistor  33 , connected between the feedback node FB and the output OUT, here formed by a plurality of resistor elements  33 ′, electrically connected in series between the feedback node FB and the output OUT; and a shifting current generator  34 , connected between the feedback node FB, on which it supplies a shifting current I t , and the reference terminal GND. 
         [0052]    The control device  30  further comprises an amplifier or amplification module  36 , having a first input that is connected to the feedback node FB and receives the feedback voltage V FB , a second input that receives a reference voltage V ref , indicative of a desired value for the output voltage V out  (for example, generated by a voltage-reference generator), and an output that supplies a control voltage V c , as a function of the difference (or error) between the feedback voltage V FB  and the aforesaid reference voltage V ref . 
         [0053]    In a manner similar to what has been indicated previously, the control voltage V c  is supplied to the PWM control unit  13 , which, starting from the same control voltage V c , generates in a suitable manner (via a voltage control mode, or a current control mode) the command signal S GD  for the switch element  4 , in such a way as to minimize the error between the reference voltage V ref  and the feedback voltage V FB . 
         [0054]    In the embodiment illustrated, the amplification module  36  includes a differential amplifier  36 ′ operating in a current mode, which generates an output current I a , according to the following expression: 
         [0000]        I   a =( V   ref   −V   FB )/ R   eq , 
         [0000]    where R eq  is the resistance of a resistor internal to the amplification module  36  (as described in greater detail hereinafter). 
         [0055]    The amplification module  36  further comprises a gain resistor  37 , having a first terminal connected to the output of the amplification module  36 , to which it supplies the output current I a  and present on which is the control voltage V c , which is consequently given by the following expression: 
         [0000]        V   c =( V   ref   −V   FB )· R   g   /R   eq  
 
         [0000]    where R g  is the resistance of the aforesaid gain resistor  37 , in the case where the second terminal of the gain resistor  37  is connected to the ground terminal of the device. 
         [0056]    In the embodiment illustrated, the control device  30  further comprises a buffer module  38 , including an operational amplifier  39  in voltage-follower configuration, connected between the output of the amplification module  36  and a second terminal of the gain resistor  37 . 
         [0057]    The presence of this buffer module  38  at the second terminal of the gain resistor  37 , instead of the connection to ground of the same terminal, advantageously allows maintaining the value of the control voltage V c  at the output of the amplification module  36  in the presence of a zero current on the gain resistor  37 , i.e., with a substantially zero error at input to the same amplification module  36 , between the reference voltage V ref  and the feedback voltage V FB , and thus to maintain a substantially zero error on the value of regulation of the output voltage V out . 
         [0058]    In greater detail, and with reference to  FIG. 7 , in a possible embodiment the shifting current generator  34  of the voltage shifter module  32  comprises an operational transconductance amplifier (OTA)  40 , having a first input terminal that receives a shifting reference voltage V LV , having an appropriate low-voltage value, operating as a voltage reference. This shifting reference voltage V LV , in a manner not illustrated but that will be evident to a person skilled in the field, may be generated in any known manner (for example, by a voltage-reference generator, of a band-gap type) within the power converter  1 D, starting from a corresponding supply voltage, or else may be received from outside by a communication bus, for example, from a control unit of an electronic apparatus where the power converter  1 D is used. The value of the shifting reference voltage V LV  may also be selected from among a set of possible references allowing to select the regulation voltage V OUT  of the entire converter. 
         [0059]    The shifting current generator  34  further comprises: a load resistor  42 , connected between a second input terminal of the operational transconductance amplifier  40  and the reference terminal GND; and a transistor element  44 , in the example of a MOSFET type, having a first current-conduction terminal connected to the feedback node FB of the regulator, a second current-conduction terminal connected to the aforesaid second input terminal of the operational transconductance amplifier  40 , and a control terminal connected to the output of the same operational transconductance amplifier  40 . 
         [0060]    The shifting current generator  34  thus generates a current: 
         [0000]    
       
      
       I 
       T 
       =V 
       LV 
       /R 
       1  
      
     
         [0000]    where R 1  is the resistance of the load resistor  42 . In a manner that will be evident, considering the entire voltage shifter circuit  32 , the feedback voltage V FB  is given by the following expression: 
         [0000]        V   FB   =V   out   −V   LV ·( R   2   /R   1 )
 
         [0000]    where R 2  is the total resistance of the shifting resistor  33 , and R 1  is the resistance of the load resistor  42 . 
         [0061]    Likewise, the output voltage V out , the regulated value of which is ensured by proper operation of the power converter  1 D, is given by the following expression: 
         [0000]        V   out   =V   FB   +V   LV ·( R   2   /R   1 )
 
         [0062]    In particular, given that the control loop operates so as to minimize the error between the reference voltage V ref  and the feedback voltage V FB , so that V ref  V FB , the aforesaid expression may be expressed as: 
         [0000]        V   out   =V   ref   +V   LV ·( R   2   /R   1 )
 
         [0063]    In other words, the value of the output voltage V out  is regulated jointly by the value of the reference voltage V ref  and by the value of the shifting reference voltage V LV . 
         [0064]    Advantageously, no attenuations from the high-frequency domain to the low-frequency domain are thus present (given that an operation of voltage shifting is, instead, envisaged). 
         [0065]    Any possible errors and noise on the low-voltage values are thus shifted towards the high voltage with unitary gain. 
         [0066]    Furthermore, the high-voltage ripple is shifted to low voltage, contributing to providing a wide input signal for the control loop. 
         [0067]    The precision of the output voltage V out  and the noise on the same output voltage V out  are moreover affected solely by the manufacturing of the voltage shifter module  32 , where, advantageously, it is possible to control in a very accurate manner the matching between the resistance values of the shifting resistor  33  and load resistor  42  (R 2 /R 1  ratio), and the value of the reference voltage V ref  and of the shifting reference voltage V LV  (depending upon which is the value of the output voltage V out ). Furthermore, the offset from the operational transconductance amplifier  40  may be optimized in an independent manner (without considering, for example, compromises of bandwidth or other characteristics with respect to other stages of the power converter  1 D). 
         [0068]    With reference to  FIG. 8 , a possible circuit implementation is now described of the amplification module  36 , and in particular of the corresponding instrumentation amplifier  36 ′, which is also referred to as a difference amplifier herein. 
         [0069]    In detail, the instrumentation amplifier  36 ′ comprises a differential input stage  46  constituted by a pair of MOSFETs  46   a ,  46   b , in the example of a PMOS type, having a respective gate terminal that receives the reference voltage V ref  and, respectively, the feedback voltage V FB . 
         [0070]    The aforementioned internal resistor, here designated by  47 , having a resistance R eq , is connected between the source terminals of the MOSFETs  46   a ,  46   b , and is traversed in use by an unbalancing current I s , which is a function of the difference (or error) between the reference voltage V ref  and the feedback voltage V FB : 
         [0000]        I   s =( V   ref   −V   FB )/ R   eq    
         [0071]    The differential amplifier  36 ′ further comprises a current-mirror stage  48 , electrically coupled to the differential input stage  46  (of a known type, here not described in detail), configured to mirror, in the example with unitary mirror factor, the unbalancing current I s  on the output of the amplification module  36 , thus generating the aforementioned output current I a . 
         [0072]    As previously mentioned, and as illustrated in  FIG. 8 , this output current I a  generates on the gain resistor  37 , connected to the output of the amplification module  36 , the control voltage V c : 
         [0000]    
       
      
       V 
       c 
       =I 
       a 
       ·R 
       g  
      
     
         [0073]    Advantageously, the solution described for the amplification module  36  is simple to implement with limited electrical consumption and area occupation in the integrated implementation. In fact, thanks to the fact that the voltage shifter module  32  does not introduce any attenuation, it is possible to use a simple single-stage gain module for the same amplification module  36 . 
         [0074]    Furthermore, the gain of the amplification module  36  may be controlled accurately, being given by a ratio between resistances, and may also be varied in a linear way (for example, in the range  1  to  30 ) by simply varying the resistance of the gain resistor  37 . 
         [0075]    The characteristics of the instrumentation amplifier  36 ′ may be optimized, for example, in terms of offset, in an independent manner, without constraints represented by the other modules of the power converter  1 D. It is, however, advantageous for the instrumentation amplifier  36 ′ to have a wider bandwidth than that of the control loop implemented by the PWM control unit  13 , in order not to introduce further singularities. Given the simple construction, also this requirement may, however, be easily satisfied. 
         [0076]    In the solution illustrated in the aforesaid  FIG. 8 , the gain resistor  37  is connected between the output of the amplification module  36  and the reference terminal GND. 
         [0077]    As described previously, and as illustrated in  FIG. 9 , the buffer module  38  may, instead, be provided, including the operational amplifier  39  in voltage follower configuration, connected between the output of the amplification module  36  and the second terminal of the gain resistor  37 . 
         [0078]    In particular, the operational amplifier  39  has a non-inverting input connected to the output of the amplification module  36 , and an inverting input, connected to the output of the operational amplifier  39 , that is connected to the aforesaid second terminal of the gain resistor  37 . 
         [0079]    The buffer module  38  acts as a further DC control loop within the control device  30 . 
         [0080]    In particular, in static steady-state conditions, i.e., when the control loop has settled the value of the output voltage V out  around the desired value, the buffer module  38  maintains the value of the control voltage V c  with an input having a zero value, i.e., with a zero regulation error (the feedback voltage V FB  is substantially equal to the reference voltage V ref ). This characteristic moreover enables a substantial immunity in regard to parasitic phenomena. 
         [0081]    In dynamic conditions, for example, in a condition of load-variation, the instrumentation amplifier  36 ′ reacts in a fast way, whereas the additional control loop defined by the buffer module  38  reacts slowly for regulating the subsequent static condition. 
         [0082]    The bandwidth of the aforesaid further control loop is thus sized in relation to the bandwidth of the main control loop for optimizing stability and performance of the regulator. Furthermore, in a manner that will be evident to a person skilled in the field, the buffer module  38  is required to absorb (or supply) the maximum current required by the instrumentation amplifier  36 ′, to prevent errors during dynamic variations. 
         [0083]    The presence of the buffer module  38  advantageously allows to maintain the value of the control voltage V c  at output from the amplification module  36 , with a zero current in the gain resistor  37 , i.e., with a substantially zero error on the input of the amplification module  36 , between the reference voltage V ref  and the feedback voltage V FB , thus maximizing the precision of regulation of the output voltage V out . 
         [0084]    The advantages of the solution proposed emerge clearly from the foregoing description. 
         [0085]    In any case, it is again emphasized that this solution provides an improved closed-loop control of a power converter  1 D, being able to ensure also a substantially zero error of regulation in static conditions (in DC conditions). The solution described may thus advantageously be used in a microelectronic system, of the type described, for example, with reference to  FIG. 10 . 
         [0086]    The microelectronic system, designated as a whole by  50 , comprises: 
         [0087]    a supply source  51 , for example, a battery supply source, which supplies a supply voltage V DD , in the low-voltage range; 
         [0088]    the power converter  1 D, for example, of a step-up type, including the control device  30 , illustrated previously in detail, designed to supply an output voltage V out , having a boosted value with respect to the supply voltage V DD  (which constitutes the input voltage V IN ); 
         [0089]    a storage element  52 , for example, a capacitor element, connected to the output out of the power converter  1 D; and 
         [0090]    a driver device  54 , which is connected to the storage element  52  and is configured to generate a driving voltage Vd, having an appropriate value in the high-voltage range, for driving, for example, electrodes of a MEMS structure  55 , which constitutes the load of the microelectronic system  50 , for example, for moving a corresponding inertial mass by the electrostatic or piezoelectric effect. 
         [0091]    Finally, it is clear that modifications and variations may be made to what is described and illustrated herein, without thereby departing from the scope of the present disclosure, as defined in the annexed claims. 
         [0092]    In particular, it is again emphasized that, even though the foregoing description has made explicit reference to a power converter  1 D of a flyback type, the present solution may advantageously be applied also to other topologies of voltage converters, not only of a step-up type (for example, boost converters), but also of a step-down type (for example, of a buck type), or of an inverter type (for example, of an inverting buck-boost type). 
         [0093]    Furthermore, as on the other hand has already been discussed previously, it is again emphasized that the solution described may be used irrespective of the control algorithm, whether with voltage or current control, implemented by the PWM control unit  13  of the power converter  1 D for generating the command signal S GD  for controlling switching of the switch element  4  of the same power converter  1 D. 
         [0094]    Finally, it is underlined that the converter according to the present solution may advantageously provide a voltage regulator or converter, to which the foregoing discussion has made explicit reference, by way of a non-limiting example, or a current regulator or converter (for example, in driver devices for LEDs or in battery chargers). The various embodiments described above can be combined to provide further embodiments. These and other changes can be made to the embodiments in light of the above-detailed description. In general, in the following claims, the terms used should not be construed to limit the claims to the specific embodiments disclosed in the specification and the claims, but should be construed to include all possible embodiments along with the full scope of equivalents to which such claims are entitled. Accordingly, the claims are not limited by the disclosure.