Abstract:
A motor current controller for a permanent magnet AC motor achieves a decrease in bandwidth variations that would otherwise be caused by temperature changes. The current controller dynamically modifies the integral gain of a proportional-integral regulator in response to temperature-induced changes in motor resistance. The integral gain value is calculated to maintain a substantially constant loop bandwidth within a current loop primarily comprising the permanent magnet AC motor, semiconductor switches connected in an H-bridge configuration, and the proportional-integral regulator.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     Not Applicable. 
     STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH 
     Not Applicable. 
     BACKGROUND OF THE INVENTION 
     The present invention relates in general to permanent magnet alternating current (PMAC) motor controls, and, more specifically, to a digital motor current controller for a permanent magnet AC motor having reduced bandwidth variations caused by temperature changes. 
     Of all the physical environments where electric motors are used, the automotive vehicle environment is one of the harshest. Operating temperatures for non-engine components can range from very cold to very hot (e.g., from about −20° C. to about +50° C.). An electric motor used in this environment must operate within its specified performance over this full temperature range. 
     Electric power assist steering systems are well known in the art. These systems often utilize a rack and pinion gear set to provide power assist by using an electric motor to either (i) apply rotary force to a steering shaft connected to a pinion gear, or (ii) apply linear force to a steering member having the rack teeth thereon. The electric motor in such systems is typically controlled in response to (i) a driver&#39;s applied torque to the vehicle steering wheel, and (ii) sensed vehicle speed. The motor may comprise a PMAC motor, a brush-type motor, or a variable reluctance motor depending upon the application and its requirements. 
     Ideally, the electric motor and control system of an electric assist steering system will have a bandwidth greater than that of the steering system itself so that the response of the electric motor does not negatively impact the stability or performance of the steering system. A substantially constant bandwidth of the motor control current loop is desirable so as to achieve consistent performance over a larger range of motor conditions, especially with regard to temperature variations. 
     Resistance of the copper windings in a permanent magnet AC motor changes with temperature. The changed response has the undesirable effect of narrowing the closed loop bandwidth of the current control loop. Therefore, the controller may require compensation for temperature changes depending upon how much impact the reduction in bandwidth has on the system performance. 
     SUMMARY OF THE INVENTION 
     The present invention has the advantages of compensating for motor resistance variations with temperature to assure a consistent response of a PMAC motor for use in a harsh temperature environment. In particular, the motor resistance variations is compensated by tuning of the motor control current loop integral gain. 
     In one aspect of the invention, a motor current controller controls an instantaneous current flow in a permanent magnet AC motor via a switching bridge in response to a current command from a main motor controller, an instantaneous current value from a motor current sensor, and a temperature signal from a motor temperature sensor. The motor current controller comprises a difference element comparing the current command and the instantaneous current value to generate an error signal. A proportional-integral regulator generates a command signal for controlling the switching bridge in response to the error signal, wherein the proportional-integral regulator includes a proportional gain value and an integral gain value. A gain control circuit generates the integral gain value and loads the integral gain value into the proportional-integral regulator. The integral gain value is determined in response to the temperature signal so that a current loop comprising the motor, the switching bridge, and the proportional-integral regulator maintains a substantially constant loop bandwidth. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic block diagram illustrating a power assist steering system in accordance with the present invention. 
     FIG. 2 is a schematic block diagram showing the digital motor current controller of FIG. 1 in greater detail. 
     FIGS. 3 a - 3   d  are plots of frequency response for comparing performance of the present invention against a motor current controller wherein the integral gain is fixed. 
    
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS 
     Referring to FIG. 1, a power assist steering system  10  includes a steering wheel  12  operatively connected to a pinion gear  14 . Specifically, the vehicle steering wheel  12  is connected to an input shaft  16  and the pinion gear  14  is connected to an output shaft  18 . The input shaft  16  is operatively coupled to the output shaft  18  through a torsion bar  20 . 
     The torsion bar  20  twists in response to applied steering torque thereby permitting relative rotation between the input shaft  16  and the output shaft  18 . Stops, not shown, limit the amount of such relative rotation between the input and output shafts in a manner known in the art. 
     As is well known in the art, pinion gear  14  has helical teeth, not shown, which are meshingly engaged with straight cut teeth, not shown, on a rack or linear steering member  22 . The pinion gear  14  in combination with the straight cut gear teeth on the rack member  22  form a rack and pinion gear set. The rack is steerably coupled to the vehicle&#39;s steerable wheels with steering linkage in a known manner. When the steering wheel  12  is turned, the rack and pinion gear set converts the rotary motion of the steering wheel  12  into linear motion of the rack  22 . When the rack moves linearly, the steerable wheels pivot about their associated steering axes and the vehicle is steered. 
     An electric assist motor  24  is operatively connected to the rack  22  through a mechanism such as a ball-nut drive arrangement, for example. When the electric motor  24  is energized, it provides power assist steering by aiding in the linear drive of the rack so as to aid in the rotation of the vehicle steering wheel  12  by the vehicle operator. 
     In accordance with the present invention, motor  24  is comprised of a PMAC motor. Motor  24  is preferably a three phase motor (which may have 9 stator slots and 6 rotor poles, commonly referred to as a “9-6” design). The stator poles are arranged so as to be energized in pairs thereby forming the 3 phases of the motor. 
     In many PMAC motors, controlling the direction of current flow through the motor windings controls direction of motor rotation. Current needs to be passed through the stator coils in only one direction independent of the desired direction of motor operation. The direction of motor rotation is controlled by the sequence in which the stator coils are energized. 
     Controlling the current through the stator coils controls the torque produced by the motor. When the assist steering motor is energized, the rotor turns which, in turn, rotates the nut portion of the ball-nut drive arrangement. When the nut rotates, the balls transfer a linear force to the rack. The direction of rack movement and, in turn, the direction of steering movement of the steerable vehicle wheels, is dependent upon the direction of rotation of the motor. 
     A steering shaft position sensor  26  is operatively connected across the steering input shaft  16  and the steering output shaft  18  and provides an electric signal having a value indicative of the relative rotational position or relative angular orientation between the input shaft  16  and the output shaft  18 . The position sensor  26  in combination with the torsion bar  20  form a torque sensor  28  that provides a torque signal having a value indicative of the applied steering torque. The output of the torque sensor  28  is connected to a torque processor  30 . Torque processor  30  monitors the applied steering torque angle, and knowing the spring constant of the torsion bar  20 , provides an electric signal indicative of the applied steering torque to a main motor controller  32 . The applied torque signal from torque processor  30  may be passed through a filtering circuit (not shown) of the type disclosed in U.S. Pat. No. 5,504,403. 
     Main motor control  32  receives a motor current signal from a current sensor  34  in additional to other input signals from other sensors (not shown) such as a motor rotor position sensor, a control system temperature sensor, and a vehicle speed sensor. Main motor control  32  provides a motor current demand signal i cmdΦ  and a phase select signal Φ sel  in response to the applied torque signal and other inputs as known in the art. The current demand signal may be determined from a look-up table in response to the measured parameters. 
     Main motor control  32  outputs the motor current demand signal i cmdΦ  and the phase select signal Φ sel  to a motor current controller  36 . A motor temperature sensor  38  is operatively coupled to PM motor  24  and provides motor current controller  36  a signal indicative of the temperature of motor  24 . The output from motor current sensor  34  also provides a motor current signal to motor current controller  36 . 
     The output of motor current controller  36  controls the current provided to each motor phase through a drive circuit  40  which is controllably connected to a plurality of power switches  42 . Power switches  42  are preferably comprised of field-effect-transistors (“FETs”) and are operatively connected between the vehicle battery B+ and the stator coils of motor  24 . Current flowing in the motor stator coils is controlled by pulse-width-modulating (“PWM”) switches  42  connected in an H-bridge configuration as is known in the art. 
     Referring to FIG. 2, motor current controller  36  is shown in greater detail. Motor current controller  36  controls current to all phases of PM motor  24  by processing the current command signal i cmdΦ  and the phase select signal Φ sel  from main motor control  32 . The purpose of motor current controller  36  is to maintain a consistent bandwidth that is independent of the motor operating parameters of rotor position, motor current, and motor temperature. 
     The current command i cmdΦ  from motor control  32  for a phase to be energized is connected to a positive input of a feedback summing junction  48 . The negative input of the summing junction  48  is connected to receive an instantaneous current measurement signal from current sensor  34 . The output of summing junction  48  is an error value Δi cmd  which is equal to the value of the difference between the reference current command value and the sensed motor current value. Error value Δi cmd  is connected to a proportional-integral (“PI”) regulator  50 . PI regulators are known in the art and are especially well adapted for controlling a PM motor in the present invention. PI controller  50  may preferably take the following form:          e   cmd     =           K   p     ·   Δ                     i   cmd       +     (           K   i     s     ·   Δ                     i   cmd       )                              
     where K p  and K i  are the proportional and integral gains, respectively. Current error command e cmd  is output from PI regulator  50  to a PWM formatter  52 . 
     The open loop transfer function of the current loop (i.e., including the motor phase windings, harness wiring, the motor current controller, and the H-bridge power switches) can be expressed as:            G        (   s   )       =       (     1     Ls   +   R       )     ·     (       K   p     +       K   i     s       )     ·     e     -     sT   d             ,                          
     where L and R are the lumped motor inductance and resistance in the DQ-frame, and T d  is the total time delay of the current loop. 
     As verified by root locus analysis, minimal variation in system bandwidth over temperature can be achieved by controlling integral gain to track changes in motor resistance. Therefore, an objective of the invention is to track the open loop zero at s=−K i /K p . In one embodiment, this is done by implementing a constant gain between the open loop zero and the open loop pole, expressed as follows:              K   i          (   T   )         K   p       =       K   ′     ·       R        (   T   )       L                              
     where K′ is the tracking gain and K i (T) and R(T) are the integral gain and motor resistance, respectively, at a temperature T. Tracking gain K′ would typically be chosen to be less than one due to the difficulty in accurately quantifying all the system parameters. The actual value used will depend on the accuracy of the various system measurements and the need for stability margins in the controller design. Preferably, K′ has a value in the range of about 0.5≦K′&lt;1. 
     The main component of the temperature-sensitive loop resistance is the motor copper resistance which is a function of temperature T as follows: 
     
       
           R ( T )= R   nom ( T   nom )+Δ R ( T ) 
       
     
     where T nom  is a nominal temperature (e.g., 20° C.), R nom (T nom ) is nominal resistance at T nom , and ΔR is the resistance change with temperature which is further given by:          Δ                   R        (   T   )         =         R   nom          (     1       T   nom     +   C       )            (     T   -     T   nom       )                              
     where C is a temperature coefficient which may be determined empirically for a particular motor. 
     Combining the above and solving for the instantaneous integral gain value to be used an a particular temperature yields:            K   i          (   T   )       =         K   i          (     T   nom     )       +       (       K   ′                       K   p     L       )        Δ                     R        (   T   )       .                                
     The nominal integral gain value K i (T nom ) can be derived conventionally for obtaining a desired bandwidth and other properties. This equation is implemented in an integral gain calculator  54  in response to a temperature signal from temperature sensor  38 . Based on the temperature signal, the integral gain value is thus calculated to maintain a substantially constant loop gain within a current loop primarily comprising the PM motor, the FET switches, and the PI regulator. In order to avoid stability problems, the integral gain value K i (T) may preferably be allowed to vary only within a predetermined range between a minimum value K i (min) and K i (max). Thus, if the above equation produces a value outside the range, then the integral gain value would be forced to the respective minimum or maximum value. Calculator  54  loads the calculated integral gain value into PI regulator  50 . 
     Using the loaded integral gain value (and preferably a constant proportional gain value programmed into regulator  50 ), the current error command e cmd  is output to PWM formatter  52 . Phase select signal Φ sel  from controller  32  is also connected to PWM formatter  52 . The output of PWM formatter  52  is controllably connected to the upper and lower switching FETs to control the PWM ON times for each of the motor phases to energize the motor in the desired direction with the desired torque. 
     A specific example of the present invention will now be described. A nominal value for inductance L of 50 μH and a nominal resistance value R of 57 mΩ will be used with a total loop delay T d  of 2.38 milliseconds and a proportional gain K p  of 50. Tracking gain K′ was selected to be 0.8 and a nominal integral gain value of 30 was chosen. FIGS. 3 a - 3   d  show results of a simulation done over a temperature range from −40° C. to +80° C. Over this temperature range, the resistance changes from 52.6 mΩ to 61.4 mΩ. FIGS. 3 a  and  3   b  show closed loop frequency response when the integral gain is not updated to track the resistance changes (i.e., K′ is zero). In this case, the current loop bandwidth ranges from 254.7 Hz to 182.3 Hz at the extremes of the temperature and resistance values. FIGS. 3 c  and  3   d  shows the frequency response when integral gain tracking (K′=0.8) is used. In this case, the bandwidth ranges from 243.2 Hz to 229.6 Hz which is a much smaller range of variation.