Abstract:
A bit sequence (b, b′) from QPSK or QAM symbols is decoded, in which an associated receive probability (w, w′) is assigned to each receive bit (b, b′). The receive probability (w, w′) is adaptively determined taking into account the transfer properties of the channel.

Description:
PRIORITY INFORMATION 
     This application claims priority from German application 10 2004 025 826.0 filed May 24, 2004, which is hereby incorporated by reference. 
     BACKGROUND OF THE INVENTION 
     The invention relates to a method and apparatus for decoding a bit sequence from QPSK or QAM symbols 
     There are generally two known fundamental techniques for decoding QPSK or QAM symbols-hard demapping and soft demapping. In the hard demapping approach, the individual received QPSK or QAM signals (received vectors) are assigned based on an unambiguous decision to a constellation point (symbol vector) (see  FIG. 16 ). With soft demapping, decoding of the received signals as data is performed, from which data and the reliability of a given decision for a specific QPSK or QAM symbol is obtained (see  FIG. 17 ). Examples of these soft-decision-output demappers are found in U.S. Pat. Nos. 6,661,282; 6,115,435; 6,226,333 and 6,424,685 as well as the article by Tosato F., Bisoglia P. “ Simplified Soft - Output - Demapper for Binary Interleaved. COFDM with Application to HIPERLAN/ 2”, Research Report, Department of Electronics, University of Padova, 2001. 
     This soft decision decoding, which weights the demodulated data by the error probability of the data, results in an improved forward error correction. For communication systems that utilize M-level QAM, the receiver thus requires a decoding algorithm that uses a two-dimensional (complex) receive signal to calculate the corresponding soft decision values as the input signals for the channel decoder. The prerequisite to ensure the reliability of this type of system is that the correct occurrence probability parameters for a given symbol are used as the basis for calculating the corresponding soft decision values. 
     As a rule, the receiver operates according to the maximum likelihood principle in which the individual probabilities are each multiplied and the receive sequence with the highest overall probability is selected. The main approach to determining the required individual probabilities is to use the Euclidean distance between receive vector and the nearest ideal symbol vector. In addition, it is generally assumed that the transfer channel shows a Gaussian amplitude distribution for the noise. Given a high signal-to-noise ratio, the logarithmic maximum likelihood function for this transfer channel is assumed to be approximately represented by:
 
LLR˜(CTF(i) 2 /σ 2 *(min[r(i)−α 0 ]−min[r(i)−α 1 ] 2 )
 
where:
     i is an index for the carrier I,   CTF is a noise amplitude of the channel transfer function,   σ 2  is a noise variance in the transfer channel,   r(i) is a receive vector with the coordinates I/Q,   α 0  is the set of constellation points that correspond to a transmitted “0” (corresponding to the “ideal” symbol vectors for a transmitted “0”), and   α 1  is the set of constellation points that correspond to a transmitted “1” (corresponding to the “ideal” symbol vectors for a transmitted “1”).   

     In coded orthogonal frequency division multiplexing (COFDM) systems, the soft information for the forward error correction (FEC) should for this reason be computed from the energy of the given carrier, the detected noise energy, and the probability of the corresponding constellation point. 
     The prior-art approach to accomplishing this starts with a fixed noise energy. 
     Calculation of the soft information is frequently implemented using mapping or lookup tables, see U.S. Pat. No. 6,115,435. Using this approach, the handling of the various constellations or hierarchy modes, such as those supporting, for example, DVB-T (Digital Video Broadcasting—terrestrial), specifically, 16-QAM, 64-QAM, a non-hierarchical constellation, a hierarchical constellation, et cetera, is difficult. 
     If, on the other hand, the decoding characteristic is calculated explicitly, the implementation is often either too complex, or significant approximation errors occur. For example, although U.S. Pat. No. 6,424,685 provides a comparatively simple calculation of the decoding characteristic from polar coordinates, considerable effort is required to adapt to the different constellations or hierarchy modes. 
     To simplify the decoding process, recent publications propose a transformation of the received constellation vectors into a simpler constellation arrangement. The term used here is “remapping”. For example, U.S. Pat. No. 6,661,282 describes a remapping by subtraction of an offset. However, this procedure is suitable only for the 16-QAM method. However, U.S. Pat. No. 6,226,333 describes the decoding of QAM symbols from a single quadrant by employing a rotator. 
     Therefore, there is a need for a technique of decoding QPSK or QAM symbols in which different constellations and hierarchy modes are easily implementable. In addition, the technique should have a high degree of reliability in predicting the decoded QPSK or QAM symbols. 
     SUMMARY OF THE INVENTION 
     According to an aspect of the invention, a bit sequence (b, b′) from QPSK or QAM symbols received following transmission over a channel is decoded, and an associated receive probability (w, w′) is assigned to each receive bit (b, b′). The receive probability (w, w′) is adaptively determined as a function of the transfer properties of the channel. 
     These and other objects, features and advantages of the present invention will become more apparent in light of the following detailed description of preferred embodiments thereof, as illustrated in the accompanying drawings. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         FIG. 1  is a block diagram illustration of a first system for decoding QPSK or QAM symbols (soft demapper); 
         FIGS. 2A-2B  show partitions of the QPSK constellation, specifically (a) MSB of the in-phase coordinate, and (b) MSB of the quadrature coordinate, respectively; 
         FIG. 3  is a plot of weighting functions for the adaptive adaptation for the system illustrated in  FIG. 1  in QPSK operation; 
         FIG. 4  is a block diagram illustration of a second system for decoding QPSK or QAM symbols (soft demapper); 
         FIGS. 5A-5D  illustrate partitions of the non-hierarchical 16-QAM constellation; 
         FIGS. 6A-6C  illustrate a rearrangement of the QAM symbols of the 16-QAM constellation (remapping) in the first quadrant of the constellation diagram,  FIG. 6A  illustrates the extraction state;  FIG. 6B  illustrates the state after shift in the Q direction by a 1 ; and  FIG. 6C  illustrates the state after shift in the I direction by b 1 ; 
         FIGS. 7A-7E  illustrate a rearrangement of the QAM symbols of the 16-QAM constellation (remapping) in the 3 rd  quadrant of the constellation diagram,  FIG. 7A  illustrates the extraction state;  FIG. 7B  illustrates the state after shift in the Q direction by a 2 ;  FIG. 7C  illustrates the state after shift in the I direction by b 2 ;  FIG. 7D  illustrates the state after reflection on the I axis; and  FIG. 7E  illustrates the state after reflection on the Q axis; 
         FIGS. 8A-8B  illustrate the log-likelihood ratios of the MSB and LSB, respectively of the I coordinate of the non-hierarchical 16-QAM constellation of  FIG. 5  (applies analogously to the MSB/LSB of the Q coordinate of the 16-QAM constellation); 
         FIGS. 9A-9B  illustrate a section of the functions of  FIGS. 8A and 8B  with Q=0, and a shift of the curve for the LSB; 
         FIG. 10  illustrates coordinates for the QAM symbols of a hierarchical 16-QAM constellation and the remapping thereof; 
         FIG. 11  is a block diagram illustration of an arithmetic unit for adaptively computing soft information; 
         FIG. 12  is a block diagram illustration of a third embodiment for decoding QPSK or QAM symbols (soft demapper); 
         FIG. 13  illustrates symbol coordinates of a non-hierarchical 64-QAM constellation; 
         FIGS. 14A-14C  illustrate the log-likelihood ratios LLR of the MSB ( FIG. 14A ), 2 nd  SB ( FIG. 14B ), and the LSB ( FIG. 14C ), of the I coordinate of the non-hierarchical 64-QAM constellation in  FIG. 13 ; 
         FIGS. 15A-15B  illustrate a section of the functions of  FIGS. 14A and 14B  where Q=0 ( FIG. 15A ) and shift of the curves for low-order bits ( FIG. 15B ); 
         FIG. 16  shows a prior art hard-output demapper; and 
         FIG. 17  shows a prior art soft-output demapper. 
     
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
       FIG. 1  is a block diagram illustration of a first decoding system  1  for decoding QPSK or QAM symbols. The decoding system  1  is suitable preferably for decoding QPSK symbols. However, the system can also be employed to decode M-level QAM symbols, for example for M=16, 64, or larger. However, to accomplish this parallel processing (not sequential signal processing, as utilized in the embodiments of  FIG. 4  et seq.) is required. 
     In the decoding system  1  includes a circuit to decode QPSK or QAM symbols by a “hard” decision (hard-decision-output demapper)  2 , a circuit  3  to determine the receive probability w for a bit, and a circuit  4  for additional weighting of the probability w by a factor G. 
     The hard-decision-output demapper  2  receives signal vectors r with coordinates I/Q on a line  100 . The hard-decision-output demapper  2  provides an output HD (hard decision) on a line  102  to tap a so-called “hard” decision b. The hard-decision-output demapper  2  provides a second output on a line  104  to the circuit  3  to determine the receive probability w of a bit. An output of this circuit  3  on a line  106  is in turn connected to the circuit  4  to weight the receive probability by a factor G. The weighting circuit  4  provides an output SD (soft decision) on a line  108  from which “soft” decision information g can be tapped. The circuits  3 ,  4  have control inputs on a line  110  through which time-variant and, in the case of multicarrier systems, carrier-dependent information about the carrier energy S, noise energy N, and/or interference can be supplied. 
     The demapping procedure is described below for a QPSK constellation: 
     An input vector r, which has in-phase coordinate I and quadrature coordinate Q and is input on the line  100  to the hard-decision demapper  2 , is assigned internally to an ideal symbol vector α and an associated bit sequence b. This bit sequence b is output from the hard-decision-output demapper  2  on the line  102 . In addition, the Euclidean distance a of the received signal vector r relative to the decision threshold  7 ,  8 , ( FIGS. 2A and 2B ) is determined within the hard-decision-output demapper  2 . This value a subsequently undergoes a soft-decision procedure. In circuit  3 , the value a on the line  104  is subjected to a demapping characteristic W, determined by the local noise energy N and/or the interference energy IF, as a function of the carrier energy S (which may be derived, for example, from a channel transfer function CTF). The result obtained by this operation is a receive probability value w output on the line  106  for the corresponding bit. 
       FIG. 3  illustrates an example of a family of curves for demapping characteristic W given different receive conditions. Whereas in response to high receive quality (high SINR) the probability w of a correct decision rises superproportionally with the increasing distance a (w 1 ), given a low-level signal to interference and noise ratio SINR, the result is instead a linear relationship between a and w (w 2 ), or in response to interferences, an actual decreasing probability w for larger distances a (w 3 ). The thus generated output signal w, as in the present embodiment, is weighted by a quantity G as a function of carrier energy S and/or noise energy N and/or interference energy IF. The output quantity thus obtained is:
   g=G*W ( a ) 
The weighting factor G which may preferably be employed here is the ratio SINR of the instantaneous signal energy S relative to the sum of the instantaneous noise and interference energies N, IF for the associated channel.
 
       FIG. 4  is a block diagram illustration of a second decoding system  41 . The decoding system  41  of  FIG. 4  comprises a circuit  42  to decode QAM symbols based on a “hard” decision (hard-decision-output demapper), a circuit  43  to determine the receive probability w for a bit, and a circuit  44  for the additional weighting of the probability w by a factor G, as well as a remapper  45  to rearrange QAM symbols. The remapper  45  for rearranging QAM symbols receives the hard-decision-output demapper  42 . 
     The decoding of a signal vector is described below using the example of a 16-QAM constellation: 
     An input vector r on the line  46  with in-phase coordinate I and quadrature coordinate Q is supplied to the remapper  45  and resolved step-by-step into sub-constellations. In a first step, input vector r is passed directly on to the hard-decision-output demapper  42 . The hard-decision-output demapper  42  makes a hard decision by assigning the receive vector r to the two most-significant bits b h  of the closest ideal symbol vector α. The soft information is determined in a procedure analogous to that described for QPSK. 
     After the initial hard decision, only a subset of possible ideal symbol vectors a remain. This remaining sub-constellation is selected in the remapper  45 . Through appropriate transformation, this sub-constellation is transformed to a constellation symmetrical with the origin. This transformation involves a shift and, as necessary, a subsequent reflection. 
     If one starts with a non-hierarchical 16-QAM constellation as found, for example, in  FIG. 5 , the result may be the transformations shown in  FIGS. 6 and 7 . 
     If one assumes that, as in  FIG. 6 , the symbol vectors a in the first quadrant are selected as the sub-constellation, the transformation comprises a shift by the shift vector a 1  and by the shift vector b 1 . 
     If one assumes that, as in  FIG. 7 , the symbol vectors a in the third quadrant are selected as the sub-constellation, the transformation comprises a shift by the shift vector a 2 , a shift by the shift vector b 2 , and two reflections c 2  and d 2 . 
     If one starts with a hierarchical 16-QAM constellation, the result may be, for example, the transformation shown in  FIG. 10 . As compared with the non-hierarchical case, only the shift vector changes here. 
     To implement the decision of the least significant bit, the receive vector r′ transformed into this constellation with in-phase coordinate I′ and quadrature coordinate Q′ is supplied to the input of the hard-decision-output demapper  42 . 
     The transformed receive vector r′ is assigned internally to a transformed ideal symbol vector α′ and to an associated bit sequence b′. This bit sequence b′ can be tapped as a hard decision at the output HD of hard-decision-output demapper  42 . 
     In addition, within the hard-decision-output demapper  42 , the Euclidean distance a′ for the now transformed received signal vector r′ is determined relative to the decision threshold  7 ,  8  used for hard decision b′—as shown in  FIG. 2 . 
     The value a′ is once again subjected to a soft-decision procedure. Referring still to  FIG. 4 , in the circuit  43 , a′ is subjected to demapping characteristic W, determined by local noise energy N and/or interference energy IF, as a function of carrier energy S. The result obtained from this operation is a receive probability value w′ for the corresponding bit. 
     The log-likelihood ratio (LLR), generally employed to determine the receive probability for a bit, is ideally a function of the in-phase coordinate I and the quadrature coordinate Q, and thus a two-dimensional function.  FIG. 8A  shows the log-likelihood ratios (LLR) for the most significant bit (MSB);  FIG. 8B  shows the log-likelihood ratios (LLR) for the least-significant bit (LSB) of a non-hierarchical 16-QAM constellation. As  FIGS. 8A and 8B  illustrate, the effect of quadrature coordinate Q for the decision in terms of in-phase coordinate I is extremely small, and may thus be neglected. As an approximation, the log-likelihood ratio (LLR) for in-phase coordinate I can be assumed to be the log-likelihood ratio with Q=0 ( FIG. 9A ). 
     The LLR characteristic for bits b of varying significance does vary considerably. However, it turns out that by appropriately shifting the individual characteristics for bits of different significance, a uniform overall characteristic is obtained which represents a sufficient approximation within the relevant control range ( FIG. 9B ). 
     This appropriate shift is implemented by the remapping procedure described above. As a result, a function that is uniform for all bits can be employed as demapping characteristic W, that is, the log-likelihood ratio (LLR) of the most significant bit MSB. 
     The characteristic W may be implemented, for example, by combining linear segments.  FIG. 11  illustrates a circuit that calculates the characteristic W using an offset correction value O and subsequent amplification value, V, on lines  1102  and  1104 , respectively. The offset correction parameter value O and the amplification parameter V are preferably stored in a table in circuit  3  provided for the demapping procedure, and are selected by arithmetic unit  6  as a function of the constellation used, the hierarchy mode used, and the adaptation parameters signal energy S, noise energy N, and interference energy IF. 
     In this embodiment, output signal w′ generated by the circuit  3  is also weighted by a quantity G dependent on the carrier energy S and/or the noise energy N and/or the interference energy IF. The output quantity obtained is:
 
 g′=G*W ( a ′)
 
The ratio SINR of the instantaneous signal energy S relative to the sum of the instantaneous noise and interference energies N, IF, of the associated channel may again be employed as the weighting factor G.
 
       FIG. 12  is a block diagram illustration of yet another circuit  1200  for decoding QPSK or QAM symbols (soft demapper). In this embodiment, a hard-decision-output demapper  1202  and a remapper  1205  are combined. Otherwise, this circuit  1200  is substantially the same as the embodiments illustrated in  FIGS. 1 and 4 . 
     The combined de-/remapper  1202 ,  1205  may, first of all, be of similar design to that of the embodiment of  FIG. 4 , that is, by connecting remapper  5  on the input side of hard-decision-output demapper  2 . This arrangement is also capable of decoding higher-level constellations, such as, for example, 64-QAM. To accomplish this, it is simply necessary to have additional iteration cycles of the above-described demapping and remapping procedure. For the sake of completeness, the diagrams corresponding to the diagrams for the non-hierarchical 16-QAM constellation of  FIGS. 5-9  are shown in  FIGS. 13-15  for the non-hierarchical 64-QAM constellation. 
     It is also possible to have the remapper  1205  connected on the output side of the demapper  1202 , or to combine the remapper  1205  and demapper  1202  in a single circuit. 
     In addition, it is possible to implement the circuit  3  for determining the receive probability w of a bit and the circuit  4  for weighting receive probability w within a single circuit. 
     The following possibilities may be considered in regard to the control quantities S, N, IF for the adaptation of the circuits  3  and/or  4 : 
     Case 1: Noise energy N is assumed to be constant; only signal energy S is determined from channel transfer function (CTF): S˜abs(CTF) 2 ; interference IF is neglected. 
     Case 2: Noise energy N is a function of carrier i; signal energy S is determined from channel transfer function CTF: S˜abs(CTF) 2 , interference IF is neglected. 
     Case 3: Noise energy N is constant; signal energy S is determined from channel transfer function CTF: S˜abs(CTF) 2 ; interference IF is determined for each carrier, or possibly estimated. 
     Case 4: noise energy N is a function of carrier i; signal energy S is determined from channel transfer function CTF: S˜abs(CTF) 2 ; interference IF is determined for each carrier, or possibly estimated. 
     Although the present invention has been shown and described with respect to several preferred embodiments thereof, various changes, omissions and additions to the form and detail thereof, may be made therein, without departing from the spirit and scope of the invention.