Abstract:
A ballast, powered by a power source is used to control operation of a load such as a discharge lamp. The ballast includes a switching network configured to control a supply of power to the load, and a bridge converter network which is configured to receive an input signal from the power source and convert it into a form useable by the switching network. The bridge converter network is integrated into the switching network.

Description:
BACKGROUND OF THE INVENTION 
     The present invention is directed to electronic lighting systems, and more particularly to an integrated bridge inverter circuit used in connection with a discharge lamp. 
     Existing single-stage high-power factor electronic ballasts designed for discharge lamps, such as integral compact fluorescent lamp applications have various drawbacks including an undesirably limited zero-voltage switching range, a high unnecessary component stress during operation and starting. Existing systems also have undesirably high crest factors and high harmonics&#39; content, which prevents product from compliance with International Electrotechnical Commission (e.g. IEC-61000-3-2) standards. 
     One existing electronic ballast which may be used for discharge lamps is shown for example, by Wong, U.S. Pat. No. 5,426,344. Wong discloses a self-oscillating high-power factor electronic ballast. The disclosed Wong circuit, as well as other existing ballasts, use input bridge circuit portions and inverter circuit portions which are distinct and separate from each other. The approach proposed in Wong et al. has a crest factor of 2.0 or higher, with high bus-voltage stresses, such as the voltage across C 3 , which in turn requires high voltage-rated transistors. A further disadvantage of this approach is a need to use of large EMI filter due to a discontinuous nature of the input current existing prior to the input bridge D 1 . The high-peak currents, which have higher high frequency current content, need to be filtered out by the input EMI filter. Therefore, yet a further disadvantage of existing ballasts such as Wong et al., is a high current stress on the switch transistors and resonant components. 
     To overcome shortcomings of existing prior art ballasts it has been deemed desirable to develop a ballast having a high power factor, low total harmonic distortion, and yet having an extended zero-voltage switching range, low cost, and low component stresses. 
     BRIEF SUMMARY OF THE INVENTION 
     A ballast, powered by a power source is used to control operation of a load such as a discharge lamp. The ballast includes a switching network configured to control a supply of power to the load, and a bridge converter network which is configured to receive an input signal from the power source and convert it into a form useable by the switching network. The bridge converter network is integrated into the switching network. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 illustrates a first embodiment of the present invention; 
     FIG. 2 sets forth various waveforms obtained by operation of the present invention; 
     FIG. 3 depicts the wave forms of FIG. 2 with an expanded time axis; and 
     FIG. 4 shows a second embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Turning to FIG. 1, illustrated is a first embodiment of a ballast lighting circuit  10  configured according to the concepts of the present invention. A half-bridge switching converter circuit including switches  11  and  12  is driven by driving circuit  13  which may be either a self-oscillating or an integrated circuit (IC) arrangement. Both such arrangements are known in the art and therefore are not discussed in detail. In FIG. 1, switches  11  and  12  are n-channel MOSFETS, with a drain of switch  11  connected to positive voltage bus  14 , the source of switch  12  connected to ground reference bus  15 , and the source of switch  11  and drain of switch  12  connected at node  16 . In alternative embodiments switches  11  and  12  may be other types of transistors or controllable switches. 
     An A.C. power source  20 , which may be a wall outlet or other power source, is connected to an electromagnetic interference (emi) filter configuration formed by emi capacitor  22  and emi inductor  24 . Power source  20  is then connected on one side to node  26  between diode  28  and diode  30  of a diode bridge also including diodes  32  and  34 . An opposite side of power source  20  is attached at node  36 , between diode  32  and diode  34  of the diode converter bridge. A main energy storage capacitor, such as electrolytic capacitor  37 , is connected across diodes  28  and  30 . 
     Node  16 , located between switch  11  and switch  12 , also serves as a connection point for one end of resonant inductor  40 , where a second end of resonant inductor  40  is attached to resonant capacitor  42 . The other end of resonant capacitor  42  is in turn connected to node  36  between diodes  32  and  34 . A further connection of resonant inductor  40  is to a discharge lamp  44 , such as a compact fluorescent lamp (CFL), a linear fluorescent lamp, a HID lamp, or other appropriate lamp type. A second end of discharge lamp  44  is attached at node  46 , which is the junction between clamping diodes  48  and  50 , whose opposite ends are connected to circuit buses  14  and  15 . Additional design of circuit  10 , includes a resonant capacitor  52  placed in parallel with diode  50 , a resonant capacitor  54  connected between nodes  26  and  46 , and a capacitor  56  connected in parallel to diode  32 . 
     A factor which differentiates circuit  10  from existing high frequency electronic ballasts, is elimination of the separation between the input circuit and the bridge circuit. For example, Wong et al. teaches an emi input filter circuit (L 1 , C 1 , C 2 ) followed by a separate distinct full-bridge circuit (D 1 ). In the present invention, diodes  28 ,  30 ,  32  and  34  are part of the bridge circuit and have also been integrated into the switching inverter circuit. Therefore, unlike existing electronic ballasts, bridge circuit  28 - 34  does not simply rectify input but is part of the inverter circuit driving switches  11 , 12 . Particularly, inverter circuit components are connected to the bridge circuit as well as the input power circuit components. 
     The topology of FIG. 1 provides several benefits including a high-power factor, which is in the range of 99%, with total harmonic distortion (THD) lowered to approximately 5%. Electronic ballast  10  therefore ensures the meeting of existing IEC standards such as IEC-61000-3-2 for harmonic distortion. Also, the crest factor obtained by circuit  10  is approximately 1.7, and it also minimizes the current stress to switches  11 ,  12  and resonant inductor  40 . Circuit  10 , permits an input current having a lower high frequency content than existing electronic ballasts, which reduces the stress on and size of emi filters  22 , 24 . The present circuit is also capable of functioning with a bus voltage lower than that used for operating existing two stage electronic ballasts. Particularly, in the present embodiment the bus voltage used is substantially the peak of the input voltage. 
     During operation of circuit  10 , diodes  48  and  50 , connected to the second end of discharge lamp  44  at node  46 , act to clamp the voltage at that end of lamp  44  to the bus voltage. Therefore, a squarewave, with a DC bias, signal is essentially created at junction  46 . The described squarewave is a high frequency A.C. waveform and is applied to capacitor  54 . This causes the current through capacitor  54  to force diode  28  and diode  34  to be alternatively conducting (assuming operation is on the positive half-cycle; on the negative half cycle, diodes  30 ,  32  will be alternatively conducting). The current from capacitor  54  becomes part of the input A.C. current, since diodes  48  and  50  are clamped to buses  14  and  15 , and the other end of capacitor  54  is connected to node  26  from input source  20 . 
     Turning to FIG. 2, and with continuing attention to FIG. 1, illustrated is waveform  60  representing the voltage across capacitor  54 . Waveform  60  is a high frequency envelope waveform, where the envelope is almost sinusoidal. This low-frequency sinusoidal waveform substantially matches A.C. power source voltage waveform  62 . The emi filters  22 ,  24  act to filter out high-frequency signal portions and this remaining portion of waveform  60  becomes part of overall input current  64 , which is substantially in-phase with input voltage waveform  62 . Thus, the current through capacitor  54  is part of the overall input current waveform  64 . 
     Turning to waveform  66  (FIG. 2) which represents the voltage across capacitor  42 , it can be seen that the waveform shape has similarities, but it is not exactly the same as waveform shape  60  across capacitor  54 . A difference between the voltage waveform  60  (capacitor  54 ) and waveform  66  (capacitor  42 ), is that the voltage across capacitor  42  has a more substantial offset from the zero crossover voltage, which indicates a voltage bias being generated across capacitor  42 . This bias across capacitor  42 , can cause distortion of the input current  64 . 
     To avoid this situation, circuit  10  implements capacitor  56 . The operation of capacitor  56  is illustrated by reviewing voltage waveform  68  (FIG. 2) which represents the voltage across capacitor  56 . Waveform  68  is substantially representative of the bus waveform minus the input voltage source waveform  62 . Since capacitor  42 , with the bias voltage, and capacitor  56 , are both connected at node  36 , the current through capacitor  56  is used to lower the current from capacitor  42  in order to reduce the current distortion in the input line current. As a result of the interaction between capacitors  42  and  56 , the sum of the current from capacitor  42 , capacitor  56  and capacitor  54  result in the preferred input current  64  of FIG.  2 . 
     A review of waveform  62  (which is the input voltage) and waveform  64  (which is the input current) illustrates that these waveforms are in-phase and of substantially the same shape, which verifies that circuit  10  has a high power factor and low THD. 
     Capacitor  52 , which is in parallel with diode  50 , provides a return path for a portion of current from lamp  44 . By this design, capacitor  52  ensures there is not an oversupply of current to the lamp, so a proper power balance between the power delivered to lamp  44  and power coming from power source  20  is obtained. This power balance is achieved by a ratio between the values of capacitor  52  and capacitor  54 . 
     As noted, the relationship between capacitor  42  and capacitor  56  is such that a canceling between these two components exist. To ensure proper canceling in desired situations, for example where the lamp current crest factor is to be less than 1.7, the following approximation may be used:            C   56     ≅     2.5   ·         V   I     ·     C   42           V     i                 n                 max       ·   ICF           ,                          
     where C 56  is capacitor  56 , V 1  is RMS of the lamp voltage, C 42  is capacitor  42 , V inmax  is line peak voltage, and ICF is the current crest factor. 
     Turning to the relationship between capacitors  42 ,  54  and  56 , the relationship between these capacitors to obtain desired output is:            (       C   42     +     C   54     +     C   56       )     =       P     i                 n           V     i                 n     2     ·   fs         ,                          
     where C 42  is capacitor  42 , C 54  is capacitor  54 , C 56  is capacitor  56 , P in  is the input power, V in   2  is the input RMS voltage squared, and fs is the switching frequency. The selection of switching frequency can indirectly determine the size of the C 42 , C 54  &amp; C 56  and transformer. Also, there should be a tradeoff consideration for the size of the resonant components VS. EMI and RFI of a product. 
     Since capacitors  42  and  54  are connected back to voltage power source  20 , the voltage across capacitors  42  and  54  are modulated by the input power source  20 , so the equivalent capacitor to the resonant tank is changed due to the modulation of the voltage across capacitor  42 . Therefore, to do the analysis or design of such a circuit as circuit  10 , it is possible to consider a modulation effect on a capacitor over the entire line cycle by using an equivalent capacitor value. 
     Over a line cycle, the effective value of capacitor on the tank circuit is changed so it becomes desirable to find what is an equivalent capacitor over the entire line cycle. The following equation finds such an equivalent capacitor for capacitor  42 :            C     42                 equ       =       2   ·     [       C   42     π     ]       +       C   42     ·   2   ·       V   I         V   I     +     V     i                 n                 max                 ,                          
     the equation for the equivalent capacitor  52  is:          C     54                 equ       =       2   ·     C   54       π                            
     A reason circuit  10  has a small amount of high-frequency current ripple in the line, compared to existing electronic ballasts, is due to the connection of one end of each of capacitor  42  and capacitor  54  to both sides of the input power source  20 , at nodes  26  and  36 . As a result of this, the currents through capacitors  42  and  54  are out of phase, which reducing peak current. Therefore, instead of having a very high peak current with a low duty ratio, the input current has lower peak current with a larger duty ratio. 
     Turning back to FIG. 2, waveforms  60 - 68  have been previously discussed. Waveform  70  is the current across switching diode  28  and waveform  72  illustrates the current waveform across switching diode  34 . As can be seen, by looking at the envelopes of these waveforms, high-frequency switching is taking place, where diodes  28  and  34  are alternatively active. Therefore, the two diodes do not conduct at the same time. This can be seen more clearly in FIG. 3, which depicts expanded timelines of waveforms including waveforms  70  and  72 . For example, in a first time period, diode  28  is not conducting during waveform portion  76  of waveform  70  while diode  34  is conducting during waveform portion  78  of waveform  72 . In a transition, diode  28  turns on at portion  80 , and diode  34  turns off at portion  82 . These changes are occurring in the conducting envelope portions  84  and  86  of FIG.  2 . 
     Lastly, with regard to FIG. 2, current waveform  88  depicts the lamp arc waveform  44 , and illustrates a low crest factor obtained without large peak current. The remaining waveforms of FIG. 3 offer expanded versions of waveforms  60 - 68  and  84 . 
     FIG. 4, sets forth a second embodiment of a ballast lighting circuit  90  of the present invention. Components which are the same as that shown in FIG. 1 are similarly numbered. In this embodiment, a change is that both capacitor  42  and capacitor  54  are tied to the same side of power source  20 . In this embodiment the connection of capacitor  54  at node  26  is removed and this end of capacitor  54  is tied to node  36 . It is to be appreciated that in a further embodiment rather than having capacitor  54  tied to node  36 , capacitor  42  may be tied to node  26  of FIG.  1 . Under this embodiment the side of capacitor  42  connected to node  36  would be removed and capacitor  42  along with capacitor  54  would be connected to node  26 . 
     In either of these arrangements, the high frequency ripple on the lines will be higher than in the first embodiment of FIG. 1 as current through capacitor  42  and capacitor  54  will be in phase. In this case, it would be required to have larger emi filters  22 ,  24  than would be used in the embodiment of FIG.  1 . 
     In one embodiment, the component values for a circuit such as described in connection with embodiments 1 and 2 of the present invention include: 
     Transistor Switch (Q 1 )  11  . . . IRF214 
     Transistor Switch (Q 2 )  12  . . . IRF214 
     Input Source (VAC)  20  . . . 120Vac 
     EMI Capacitor (C 6 )  22  . . . 0.22uF 
     EMI Inductor (L 2 )  24  . . . 2mH 
     Bridge Diodes (D 1 -D 4 )  28 - 34  . . . 1N4937 
     Energy Storage Capacitor (C 1 )  37  . . . 47uF 
     First Inductor (L 1 )  40  . . . 0.7mH 
     Divider Capacitor (C 3 )  42  . . . 3.3nF 
     Bridge Diodes (D 5 -D 6 )  48 - 50  . . . 1N4937 
     Parallel Capacitor (C 2 )  52  . . . 1nF 
     Capacitor (C 5 )  54  . . . 16.8nF 
     Parallel Capacitor (C 4 )  56  . . .1.3nF 
     While the invention has been described with reference to a preferred embodiment, it will be understood by those skilled in the art that various changes may be made and equivalents may be substituted for elements thereof without departing from the scope of the invention. In addition, many modifications may be made to adapt a particular situation or material to the teachings of the invention without departing from the essential scope thereof. Therefore, it is intended that the invention not be limited to the particular embodiment disclosed as the best mode contemplated for carrying out this invention, but that the invention will include all embodiments falling within the scope of the appended claims.