Abstract:
This invention generally relates to wired and wireless ultra wideband (UWB) data communications apparatus and methods, and in particular to UWB receiver systems and architectures. 
     An ultra wideband (UWB) receiver system comprising: a receiver front end to receive a UWB signal; an analogue-to-digital converter coupled to said receiver front end to digitize said received UBW signal; and a correlator coupled to said analogue-to-digital converter to correlate said digitized UWB signal with a reference signal.

Description:
FIELD OF THE INVENTION 
   This invention generally relates to wired and wireless ultra wideband (UWB) data communications apparatus and methods, and in particular to UWB receiver systems and architectures. The benefit of U.S. provisional 60/518,344 filed Nov. 10, 2003 is claimed. 

   BACKGROUND TO THE INVENTION 
   Techniques for UWB communication developed from radar and other military applications, and pioneering work was carried out by Dr G. F. Ross, as described in U.S. Pat. No. 3,728,632. Ultra-wideband communications systems employ very short pulses of electromagnetic radiation (impulses) with short rise and fall times, resulting in a spectrum with a very wide bandwidth. Some systems employ direct excitation of an antenna with such a pulse which then radiates with its characteristic impulse or step response (depending upon the excitation). Such systems are referred to as carrierless or “carrier free” since the resulting rf emission lacks any well-defined carrier frequency. However other UWB systems radiate one or a few cycles of a high frequency carrier and thus it is possible to define a meaningful centre frequency and/or phase despite the large signal bandwidth. The US Federal Communications Commission (FCC) defines UWB as a —10 dB bandwidth of at least 25% of a centre (or average) frequency or a bandwidth of at least 1.5 GHz; the US DARPA definition is similar but refers to a —20 dB bandwidth. Such formal definitions are useful and clearly differentiates UWB systems from conventional narrow and wideband systems but the techniques described in this specification are not limited to systems falling within this precise definition and may be employed with similar systems employing very short pulses of electromagnetic radiation. 
   UWB communications systems have a number of advantages over conventional systems. Broadly speaking, the very large bandwidth facilitates very high data rate communications and since pulses of radiation are employed the average transmit power (and also power consumption) may be kept low even though the power in each pulse may be relatively large. Also, since the power in each pulse is spread over a large bandwidth the power per unit frequency may be very low indeed, allowing UWB systems to coexist with other spectrum users and, in military applications, providing a low probability of intercept. The short pulses also make UWB communications systems relatively unsusceptible to multipath effects since multiple reflections can in general be resolved. Finally UWB systems lend themselves to a substantially all digital implementation, with consequent cost savings and other advantages. 
     FIG. 1   a  shows an example of an analogue UWB transceiver  100 . This comprises an transmit/receive antenna  102  with a characteristic impulse response indicated by bandpass filter (BPF)  104  (although in some instances a bandpass filter may be explicitly included), couples to a transmit/receive switch  106 . 
   The transmit chain comprises an impulse generator  108  modulatable by a baseband transmit data input  110 , and an antenna driver  112 . The driver may be omitted since only a small output voltage swing is generally required. One of a number of modulation techniques may be employed, typically either OOK (on-off keying i.e. transmitting or not transmitting a pulse), M-ary amplitude shift keying (pulse amplitude modulation), or PPM (pulse position modulation i.e. dithering the pulse position). Typically the transmitted pulse has a duration of &lt;Ins and may have a bandwidth of the order of gigahertz. 
   The receive chain typically comprises a low noise amplifier (LNA) and automatic gain control (AGC) stage  114  followed by a correlator or matched filter (MF)  116 , matched to the received pulse shape so that it outputs an impulse when presented with rf energy having the correct (matching) pulse shape. The output of MF  116  is generally digitised by an analogue-to-digital converter (ADC)  118  and then presented to a (digital or software-based) variable gain threshold circuit  120 , the output of which comprises the received data. The skilled person will understand that forward error correction (FEC) such as block error coding and other baseband processing may also be employed, but such techniques are well-known and conventional and hence these is omitted for clarity. 
     FIG. 1   b  shows one example of a carrier-based UWB transmitter  122 . A similar transmitter is described in more detail in U.S. Pat. No. 6,026,125. This form of transmitter allows the UWB transmission centre frequency and bandwidth to be controlled and, because it is carrier-based, allows the use of frequency and phase as well as amplitude and position modulation. Thus, for example, QAM (quadrature amplitude modulation) or M-ary PSK (phase shift keying) may be employed. 
   Referring to  FIG. 1   b , an oscillator  124  generates a high frequency carrier which is gated by a mixer  126  which, in effect, acts as a high speed switch. A second input to the mixer is provided by an impulse generator  128 , filtered by an (optional) bandpass filter  130 . The amplitude of the filtered impulse determines the time for which the mixer diodes are forward biased and hence the effective pulse width and bandwidth of the UWB signal at the output of the mixer. The bandwidth of the UWB signal is similarly also determined by the bandwidth of filter  130 . The centre frequency and instantaneous phase of the UWB signal is determined by oscillator  124 , and may be modulated by a data input  132 . An example of a transmitter with a centre frequency of 1.5 GHz and a bandwidth of 400 MHz is described in U.S. Pat. No. 6,026,125. Pulse to pulse coherency can be achieved by phase locking the impulse generator to the oscillator. 
   The output of mixer  126  is processed by a bandpass filter  134  to reject out-of-band frequencies and undesirable mixer products, optionally attenuated by a digitally controlled rf attenuator  136  to allow additional amplitude modulation, and then passed to a wideband power amplifier  138  such as a MMIC (monolithic microwave integrated circuit), and transmit antenna  140 . The power amplifier may be gated on and off in synchrony with the impulses from generator  128 , as described in U.S. Pat. No. &#39;125, to reduce power consumption. 
     FIG. 1   c  shows a similar transmitter to that of  FIG. 1   b , in which like elements have like reference numerals. The transmitter of  FIG. 1   c  is, broadly speaking, a special case of the transmitter of  FIG. 1   b  in which the oscillator frequency has been set to zero. The output of oscillator  124  of  FIG. 1   b  is effectively a dc level which serves to keep mixer  126  always on, so these elements are omitted (and the impulse generator or its output is modulated). 
     FIG. 1   d  shows an alternative carrier-based UWB transmitter  142 , also described in U.S. Pat. No. 6,026,125. Again like elements to those of  FIG. 1   b  are shown by like reference numerals. 
   In the arrangement of  FIG. 1   d  a time gating circuit  144  gates the output of oscillator  124  under control of a timing signal  146 . The pulse width of this timing signal determines the instantaneous UWB signal bandwidth. Thus the transmitted signal UWB bandwidth may be adjusted by adjusting the width of this pulse. 
   Ultra-wideband receivers suitable for use with the UWB transmitters of  FIGS. 1   b  to  1   d  are described in U.S. Pat. No. 5,901,172. These receivers use tunnel diode-based detectors to enable single pulse detection at high speeds (several megabits per second) with reduced vulnerability to in-band interference. Broadly speaking a tunnel diode is switched between active and inactive modes, charge stored in the diode being discharged during its inactive mode. The tunnel diode acts, in effect, as a time-gated matched filter, and the correlation operation is synchronised to the incoming pulses. 
     FIG. 1   e  shows another example of a known UWB transmitter  148 , described in U.S. Pat. No. 6,304,623. In  FIG. 1   e  a pulser  150  generates an rf pulse for transmission by antenna  152  under control of a timing signal  154  provided by a precision timing generator  156 , itself controlled by a stable timebase  158 . A code generator  160  receives a reference clock from the timing generator and provides pseudo-random time offset commands to the timing generator for dithering the transmitter pulse positions. This has the effect of spreading and flattening the comb-like spectrum which would otherwise be produced by regular, narrow pulses (in some systems amplitude modulation may be employed for a similar effect). 
     FIG. 1   f  shows a corresponding receiver  162 , also described in U.S. Pat. No. &#39;623. This uses a similar timing generator  164 , timebase  166  and code generator  168  (generating the same pseudo-random sequence), but the timebase  166  is locked to the received signal by a tracking loop filter  170 . The timing signal output of timing generator  164  drives a template generator  172  which outputs a template signal and a correlator/sampler  176  and accumulator  178  samples and correlates the received signal with the template, integrating over an aperture time of the correlator to produce an output which is sampled at the end of an integration cycle by a detector  180  to determine whether a one or a zero has been received. 
     FIG. 1   g  shows a UWB transceiver  182  employing spread spectrum-type coding techniques. A transceiver of the general type is described in more detail in U.S. Pat. No. 6,400,754, to which reference may be made. 
   In  FIG. 1   g  a receive antenna  184  and low noise amplifier  186  provide one input to a time-integrating correlator  188 . A second input to the correlator is provided by a code sequence generator  190  which generates a spread spectrum-type code such as a Kasami code, that is a code with a high auto-correlation coefficient from a family of codes with low cross-correlation coefficients. Correlator  188  multiplies the analogue input signal by the reference code and integrates over a code sequence period and may comprise a matched filter with a plurality of phases representing different time alignments of the input signal and reference code. The correlator output is digitised by analogue-to-digital converter  192  which provides an output to a bus  194  controlled by a processor  196  with memory  198  the code sequence generator  190  is driven by a crystal oscillator driven clock  200  a transmit antenna driver  202  receives data from bus  194  which is multiplied by a code sequence from generator  190  and transmitted from transmit antenna  204 . In operation coded sequences of impulse doublets are received and transmitted, in one arrangement each bit comprising a 1023-chip sequence of 10 ns chips, thus having a duration of 10 μs and providing 30 dB processing gain. Shorter spreading sequences and/or faster clocks may be employed for higher bit rates. 
   The transceiver described in U.S. Pat. No. 6,400,754 uses a modification of a frequency-independent current-mode shielded loop antenna (described in U.S. Pat. No. 4,506,267) comprising a flat rectangular conducting plate. This antenna is referred to as a large-current radiator (LCR) antenna and when driven by a current it radiates outwards on the surface of the plate. 
     FIG. 1   h  shows a driver circuit  206  for such an LCR transmit antenna  208 . The antenna is driven by an H-bridge comprising four MOSFETs  210  controlled by left (L) and right (R) control lines  212 ,  214 . By toggling line  214  high then low whilst maintaining line  214  low an impulse doublet (that is a pair of impulses of opposite polarity) of a first polarity is transmitted, and by toggling line  212  high then low whilst holding line  214  low an impulse doublet of opposite polarity is radiated. The antenna only radiates whilst the current through it changes, and transmits a single gaussian impulse on each transition. 
     FIGS. 2   a  to  2   h  show some examples of UWB waveforms,  FIG. 2   a  shows a typical output waveform of a UWB impulse transmitter, and  FIG. 1   b  shows the power spectrum of the waveform of  FIG. 2   a .  FIG. 2   c  shows a wavelet pulse (which when shortened becomes a monocycle) such as might be radiated from one of the transmitters of  FIGS. 1   b  to  1   d .  FIG. 2   d  shows the power spectrum of  FIG. 2   c .  FIG. 2   e  shows an impulse doublet and  FIG. 2   f  the power spectrum of the doublet of  FIG. 2   e . It can be seen that the spectrum of  FIG. 2   f  comprises a comb with a spacing (in frequency) determined by the spacing (in time) of the impulses of the doublet and an overall bandwidth determined by the width of each impulse. It can also be appreciated from  FIGS. 2   e  and  2   f  that dithering the pulse positions will tend to reduce the nulls of the comb spectrum.  FIG. 2   g  shows examples of basis impulse doublet waveforms for a logic 0 and a logic 1.  FIG. 2   h  shows an example of a TDMA UWB transmission such as might be radiated from the transceiver of  FIG. 1   g , in which bursts of Code Division Multiple access (CDMA)-encoded data are separated by periods of non-transmission to allow access by other devices. 
   Ultra wideband communications potentially offer significant advantages for wireless home networking, particularly broadband networking for audio and video entertainment devices, because of the very high data rates which are possible with UWB systems. However, UWB communications also present a number of special problems, most particularly the very low transmit power output imposed by the relevant regulatory authorities, in the US the FCC. Thus the maximum permitted power output is presently below the acceptable noise floor for unintentional emitters so that a UWB signal effectively appears merely as noise to a conventional receiver. This low power output limits the effective range of UWB communications and there is therefore a need to address this difficulty. 
   One way to improve the range of a UWB communications link is to adopt a rake receiver type approach to combine the energy in a plurality of multipath components of a received signal. Multipath effects arise when a signal from a transmitter to a receiver takes two or more different paths (multipaths) such as a direct path between a transmit and receive antenna and an indirect path via reflection off a surface. In a multipath environment two or more versions of a transmitted signal arrive at the receiver at different times. Most wireless environments, and in particular indoor environments, have significant levels of multipath which, in a conventional RF communications system, typically produces a comb-like frequency response, the multiple delays of the multipath components of the received signal giving the appearance of tines of a rake. The number and position of multipath channels generally changes over time, particularly when one or both of the transmitter and receiver is moving. 
   It is helpful to briefly review the operation of a conventional rake receiver before going on to consider a known UWB rake-type receiver. 
   In a spread spectrum communication system a baseband signal is spread by mixing it with a pseudorandom spreading sequence of a much higher bit rate (referred to as the chip rate) before modulating the rf carrier. At the receiver the baseband signal is recovered by feeding the received signal and the pseudorandom spreading sequence into a correlator and allowing one to slip past the other until a lock is obtained. Once code lock has been obtained, it is maintained by means of a code tracking loop such as an early-late tracking loop which detects when the input signal is early or late with respect to the spreading sequence and compensates for the change. Alternatively a matched filter may be employed for despreading and synchronisation. 
   Such a system is described as code division multiplexed as the baseband signal can only be recovered if the initial pseudorandom spreading sequence is known. A spread spectrum communication system allows many transmitters with different spreading sequences all to use the same part of the rf spectrum, a receiver “tuning” to the desired signal by selecting the appropriate spreading sequence (CDMA—code division multiple access). 
   One advantage of conventional spread spectrum systems is that they are relatively insensitive to multipath fading. A correlator in a spread spectrum receiver will tend to lock onto one of the multipath components, normally the direct signal which is the strongest. However a plurality of correlator may be provided to allow the spread spectrum receiver to lock onto a corresponding plurality of separate multipath components of the received signal. Such a spread spectrum receiver is known as a rake receiver and the elements of the receiver comprising the correlator are often referred to as “fingers” of the rake receiver. The separate outputs from each finger of the rake receiver are combined to provide an improved signal to noise ratio (or bit error rate) generally either by weighting each output equally or by estimating weights which maximise the signal to noise ratio of the combined output (“Maximal Ratio Combining”—MRC). 
     FIG. 3   a  shows the main components of a typical rake receiver  300 , A bank of correlators  302  comprises, in this example, three correlators  302 ,  302  and  302  each of which receives a CDMA spread spectrum signal from input  304 . The correlators are known as the fingers of the rake; in the illustrated example the rake has three fingers. The CDMA signal may be at baseband or at IF (Intermediate Frequency). Each correlator locks to a separate multipath component which is delayed by at least one chip with respect to the other multipath components. More or fewer correlators can be provided according to a quality-cost/complexity trade off. The despread output from a correlator is a signal with a magnitude and phase modified by the attenuation and phase shift of the multipath channel through which the multipath component locked onto by the finger of the rake receiver has been transmitted. A channel estimate comprising a complex number characterising the phase and attenuation of the communications channel, in particular for the multipath component of the channel the rake finger has despread, may be obtained, for example using a training sequence. The channel estimate may then be conjugated to invert the phase (and optionally normalised) and used to multiply the received signal to compensate for the channel. 
   The outputs of all the correlators go to a combiner  306  such as an MRC combiner, which adds the outputs in a weighted sum, generally giving greater weight to the stronger signals. The weighting may be determined based upon signal strength before or after correlation, according to conventional algorithms. The combined signal is then fed to a discriminator  308  which makes a decision as to whether a bit is a 1 or a 0 and provides a baseband output. The discriminator may include additional filtering, integration or other processing. The rake receiver may be implemented in either hardware or software or a mixture of both. 
   The effects of multipath propagation on UWB transmissions are not the same as on conventional RF transmissions. In particular where a UWB signal comprises a succession of wavelets or pulses (the terms are used substantially synonymously in the specification), because of the short duration and relatively long separation (in time) of these pulses it is often possible to substantially time-resolve the pulses belonging to multipath components of the UWB signal. In simple terms, the delays between the arrival of pulses in different multipath components originating from a single transmitted UWB pulse are often long enough to make it unlikely that two pulses arrive at the same time. This is described further below and can be exploited to advantage in a UWB receiver design. 
   It is known to apply conventional rake receiver techniques to UWB communications systems, as described for example in WO01/93441, WO01/93442, and WO01/93482.  FIG. 3   b , which is taken from WO01/93482, shows such a transceiver; similar arrangements are described in the other two specifications. 
   Referring to  FIG. 3   b , this shows a UWB transmitter  7   0 ,  21 ,  17 ,  23 ,  25 ,  27 ,  1  and a UWB receiver  1 ,  27 ,  3 ,  29 ,  31 ,  1 - N ,  7   1 - N ,  9 . The receiver comprises a plurality of tracking correlators  31   1 - 31   N  together with a plurality of timing generators  7   1 - 7   N , and as described in WO &#39;482 (page 15) during a receive mode of operation the multiple arms can resolve and lock onto different multipath components of a signal. By coherent addition of the energy from these different multipath signal components the received signal to noise ratio may be improved. However the design of &#39;482 is relatively physically large, expensive and power hungry to implement and fails to take advantage of some aspects of UWB multipath transmission. 
   SUMMARY OF THE INVENTION 
   In one aspect the invention provides an ultra wideband (UWB) receiver system comprising a receiver front end to receive a UWB signal; an analogue-to-digital converter coupled to said receiver front end to digitise said received UWB signal; and a correlator coupled to said analogue-to-digital converter to correlate said digitised UWB signal with a reference signal. 
   Digitising the received UWB signal prior to correlation with a reference signal helps to reduce the cost, size, and power consumption of the receiver by facilitating further processing in accordance with a number of digital techniques as described below. For example one digital correlator correlating a pair of signals (the received and reference signals) may be employed to correlate a received pulse having a plurality of multipath components, rather than a separate correlation being needed for each separate multipath component. Correlation speed may also be increased by correlating a set of digitised samples of an incoming UWB signal with a reference signal. More generally digitising prior to the correlator facilitates time multiplexing of a correlator, for example storing partial or intermediate correlation results, and thus facilitates the provision of a much larger number of correlators than might otherwise be possible, thus helping to provide an improved signal to noise ratio. Multiple correlators may be employed in parallel, for example for pipelined correlation operations, and by multiplexing correlators in the time domain, sharing a single correlator between different multipath components of the same pulse and/or between different pulses a plurality of logically parallel correlators may be provided by a single physical correlator. In a preferred embodiment the UWB signal is a carrierless signal comprising a plurality of pulses or wavelets, thus facilitating such time multiplexing since much of the reference wave form is then substantially zero, comprising short pulses separated by longer periods of substantially zero signal, providing spaces in which a correlator or multiple-accumulator may be multiplexed for use in correlating other (later or earlier) transmitted pulses. Thus the correlator can be faster, more efficient, smaller and potentially cheaper. 
   As UWB signals by their nature have a very wide bandwidth, for example 500 MHz or greater even when the UWB spectrum is subdivided into bands, the analogue-to-digital converter preferably operates at 10 9  samples per second and may operate at 10 10  samples per second or more. Since such fast analogue-to-digital conversion can have a high power consumption the reference signal, for example from a reference signal generator, may be used to control a power supply to the converter so that the power consumption of the converter can be reduced in periods where there is no expected received signal (pulse). Thus the converter may be inhibited or its power effectively removed during periods when no multipath component of a pulse is expected. Thus in another aspect the invention provides a method of controlling an analogue-to-digital converter in accordance with these techniques. 
   As the received UWB signal may be at a very low level or even in the noise (because there may be redundancy in the transmitted data, for example by distributing a data bit over a plurality of successively transmitted pulses) an analogue-to-digital converter with a multiple bit resolution is indicated. However the number of bits required may be reduced by employing a controllable gain device such as a switched or programmeable attenuator prior to the analogue-to-digital converter and controlling this using the reference signal. Thus when a low level multipath component is expected the gain may be increased or attenuation reduced. In some circumstances a one bit analogue-to-digital converter may even be employed, which simplifies later correlation since a one-bit multiply may be implemented by means of an EXOR gate. Optionally one or more statistical characteristics of the received UWB signal may be monitored in order to adjust the signal prior to digitisation. 
   Where the UWB signal comprises a plurality of pulses a plurality of correlation or multiply-accumulate modules may be employed to perform a plurality of correlations substantially in parallel between the digitised UWB signal and differently delayed versions of the reference signal to locate a pulse. As described further below in embodiments it is highly advantageous to delay the reference signal rather than the digitised sample data as using a look-up table in memory it is relatively straightforward to generate differently delayed versions of a stored signal. In embodiments this provides much faster signal acquisition and tracking than a conventional sliding window or early-late approach thus facilitating managing the very fast stream of data from the analogue-to-digital converter. The skilled person will appreciate that the bandwidth of the digital ‘pipe’ from this converter is roughly three orders of magnitude wider than the level at which conventional digital RF processing generally takes place (10 GHz as compared with 10 MHz). 
   In preferred embodiments each of the correlation modules correlates a plurality of multipath components of a transmitted pulse and where these interleave but do not substantially overlap a correlation module may be time multiplexed to correlate multipath components of successively transmitted pulses with the reference signal. This further facilitates rapid processing of the digitised incoming UWB signal. Where correlation is performed against a plurality of multipath components a single template may be used for all multipath components but preferably a separate template is used for at least some of or each of the multipath components for improved correlation with the received signal. The reference signal template for one or more multipath components may be derived by training the receiver. For example an associated UWB transmitter may be employed to directly transmit a signal to the receiver to calibrate out or at least compensate for phase and/or gain distortions or non-linearities introduced by the receiver (analogue) front end and/or analogue-to-digital converter. This is important in a UWB system since the ultra wideband input stage will generally have phase and gain characteristics which vary significantly over the receiver bandwidth thus introducing significant distortions to the transmitted pulse shape. The associated transmitter may provide a timing signal directly to the receiver, for example across a direct wired connection, so that the receiver knows when to store received data into a reference signal memory in order to construct a receive signal template. A second training stage may optionally (but preferably) also be applied in which a signal from a remote transmitter is acquired using the stored reference signal and then used to store a second reference signal which takes account of the distortions introduced by the channel for each multipath component. 
   It will be appreciated that the above described correlator may be employed to locate the position of a pulse in time (pulse-position modulation). However in preferred embodiments the correlator includes a discriminator which may be employed to determine the phase of a received pulse, for example determining whether the pulse is upright or inverted (bi-phase modulation). This bi-phase determination effectively comes at no extra cost to the pulse position determination and is effectively co-determined with the pulse position. This permits a doubling in the number of data bits to be sent as compared with either pulse position modulation or bi-phase modulation when used separately. Although it is known to dither the position of bi-phase pulses, for example using a pseudorandom sequence, it has not previously been recognised that a pulsed UWB signal may be modulated with information data to be communicated across a UWB link using pulse position modulation (PPM) in which a pulse position is dependant upon the information data and at the same time bi-phase modulation (or M-ary phase modulation) in which the pulse phase is also dependant upon the information data, two or more information bits being encoded as a combination of pulse phase and pulse position. 
   Preferably the analogue-to-digital converter comprises a plurality of A-to-D conversion modules to provide a plurality of successively delayed samples of the UWB signal substantially in parallel, for example using a phase-tapped clock. The correlator may then be configured to process these successively delayed samples substantially simultaneously to allow a time-slice of the received signal to be correlated with a time-slice of the reference signal, preferably the time-slice having sufficient duration to encompass a substantial portion of a pulse within the UWB signal. Since a single transmitted pulse is generally received as a set of multipath components each comprising a version of the pulse the time-slice preferably extends over the duration of a pulse of a multipath component rather than over the entire set of multipath components which may be significantly delayed with respect to one another. A correlator to correlate time-slices of the received and reference signals each comprising a plurality of samples may comprise a plurality of multipliers in parallel coupled to a common accumulator. 
   The reference signal is preferably provided from a reference signal memory, and advantageously within this memory the reference signal may be stored as a set of pulse shapes each separated by a delay, each pulse shape representing an expected pulse shape of a received multipath component of a transmitted pulse. Thus in a preferred embodiment the reference signal memory stores sets of data samples defining pulse shapes and associated data values defining delays between these pulses, that is between the reception of different multipath components of a pulse. 
   Preferably the reference signal memory is configured to provide a plurality of stored reference signal values for different, successively delayed versions of the reference signal substantially simultaneously. This facilitates rapid location of the exact position in time of a multipath component of a pulse since the successively delayed versions of the pulse may be correlated with the incoming received signal in parallel. Thus the reference signal memory preferably has a plurality of outputs so that when the memory is addressed each of these outputs can provide a data sample corresponding to a different point on the stored reference signal waveform. In fact, as described above, it is preferable for the reference signal memory to have a plurality of sets of outputs each set of outputs providing a time-slice of samples of the stored reference signal extending over a period approximately equal to or greater than the duration of a multipath component of a pulse, the sets of samples being successively delayed to provide, in effect, successively delayed pulse shapes. It is further preferable that the delay of each set of samples, that is of each pulse shape, should correspond to an integral number (such as one) sample periods so that these successively delayed sets of samples may be provided by appropriate wiring of outputs from the memory. For example first and second sets of 16 samples delayed by one sample will have 15 sample values in common and need only 17 outputs from the memory to provide 32 sets of samples defining two pulse shapes delayed by one sample with respect to one another. 
   Preferably the receiver system also includes a controller for identifying data in a received UWB signal which can be written into the reference signal memory for use as a template in later correlations. This facilitates training of the receiver to perform correlations with unexpected received UWB signal rather than an ideal UWB signal. It will be appreciated that the controller need not itself read the digitised signal to be used as a template but need only identify which portion of an incoming signal may be employed for this purpose, for example using the correlator. In some embodiments more than one reference signal may be stored. For example a reference signal may be stored for each of a plurality of transmitters from which the receiver system receives signals, allowing a different reference signal to be employed for each transmitter, and therefore allowing the receiver to compensate for the potentially different multipath channel to each transmitter. The reference signal for a transmitter may then be selected according to which transmitter the receiver is communicating with (or in some embodiments correlations with more than one reference signal may be performed effectively in parallel, for example by multiplexing). 
   In a related aspect the invention provides an ultra wideband (UWB) receiver system for reception of UWB signals from a plurality of UWB transmitters, the receiver system comprising: a receiver front end to receive a said UWB signal; a correlator coupled to said receiver front end to correlate said UWB signal with a reference signal; and means to select one of a plurality of said reference signals for receiving a signal from a corresponding one of said transmitters. 
   The above described features and aspects of the invention may advantageously be combined and permuted, as will be understood by the skilled person. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     These and other aspects of the present invention will now be further described, by way of example only, with reference to the accompanying figures in which: 
       FIGS. 1   a  to  1   h  show, respectively, a typical UWB transceiver, a first example of a carrier-based UWB transmitter, a variant of this first example transmitter, a second example of a carrier-based UWB transmitter, a third example of a UWB transmitter, a receiver for the third example transmitter, a UWB transceiver employing spread spectrum techniques, and a driver circuit for a large-current radiator antenna; 
       FIGS. 2   a  to  2   h  show examples of UWB waveforms; 
       FIGS. 3   a  and  3   b  show, respectively, the main elements of a conventional rake receiver for spread-spectrum signals, and a block diagram of a known UWB transceiver employing conventional rake receiver techniques; 
       FIGS. 4   a  to  4   d  show, respectively, a transmitted UWB signal comprising a single pulse, an example of a received version of the transmitted pulse of  FIG. 4   a  with multipath reflections and other propagation effects, a series of transmitted UWB pulses of the type shown in  FIG. 4   a , and a received signal corresponding to the transmitted signal of  FIG. 4   c  showing overlapping multipath reflections; 
       FIG. 5  shows an overview block diagram of a UWB receiver embodying aspects of the present invention; 
       FIG. 6  shows a simplified block diagram of a demodulator architecture for use with the receiver of  FIG. 5 ; 
       FIG. 7  shows a timing diagram illustrating timing variations of multipath components of a pulse with respect to pulse repetition frequency; 
       FIG. 8  shows diagrammatically a modulation scheme for use with the D modulator of  FIG. 6 ; 
       FIGS. 9   a  and  9   b  show, respectively, a data frame format and pilot tone pulses for the receiver of  FIG. 5 ; 
       FIGS. 10   a  and  10   b  show, respectively, a UWB transmitter and a pulse generator for the UWB transmitter; 
       FIGS. 11   a  and  11   b  show, respectively, a signal acquisition and tracking system for the receiver of  FIG. 5 , and a waveform memory data format; 
       FIGS. 12   a  and  12   b  show, respectively, a flow diagram of a signal acquisition procedure, and a diagrammatic illustration of a signal hunt process; 
       FIGS. 13   a  and  13   b  show, respectively, a reference waveform generation system, and a variant of the system of  FIG. 13   a;    
       FIG. 14  shows a block diagram of a correlator for the demodulator of  FIG. 6 ; 
       FIGS. 15   a  and  15   b  show, respectively, received signals with interleaved multipath components, and a diagrammatic illustration of the operation of the correlator of  FIG. 14 . 
   

   DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   As previously mentioned a transmission medium coupling a UWB transmitter and UWB receiver will typically give rise to a number of physical effects that complicate the function of the receiver. The transmission medium may comprise a wireless or wired transmission channel. The physical effects include multiple path reflections, which result in multiple pulses at the receiver or each transmitted pulse, in some cases these pulses being phase inverted. Dispersion, frequency dependent continuation and other properties of the transmission medium distort the pulse shape. Interference and noise sources are received in addition to the desired pulse data. Noise sources include thermal noise (from the receiver itself), narrow band interference from radio transmitters sharing the same frequency spectrum, and broadband interference (from switching and alike). There may also be interference from co-located UWB systems sharing the same physical space for electrical cabling. A UWB receiver should preferably be capable of dealing with all these effects. 
   Referring now to  FIG. 4 ,  FIG. 4   a  shows an example of a transmitted UWB pulse, which in this example has a duration of approximately 100 ps.  FIG. 4   b  shows the same pulse as it might be seen by a UWB receiver. As can be seen the received pulse has a plurality of multipath components and also exhibits distortion and other propagation effects. Multipath components are received over a time scale which depends upon the transmission channel but which may, for example, be between 10 ns and 100 ns (the pulses shown in this diagram are not to scale), multipath at the longer end of this range being observed in wired systems such as UWB over mains (AC power cable) transmissions as described in the co-depending UK Patent Application No. 0222828.6 filed on 2 Oct. 2002. The first received multipath component need not be the largest (as shown in  FIG. 4   b ) and may be significantly distorted or even inverted. 
     FIG. 4   c  illustrates a series of transmitted pulses and  FIG. 4   d  an example of a corresponding received signal. It can be seen that multipath reflections from one pulse may overlap with the first signals from the next pulse. This problem is exasperated when timing modulation is applied to a transmitted pulse. 
     FIG. 5  shows a block diagram of a UWB receiver  500  embodying an aspect of the present invention. 
   An incoming UWB signal is received by an antenna  502 , which may comprise a capacitive an/or inductive coupling to a cable system such as a mains power cable or a telephone cable. The received UWB signal is provided to an analog front end block  504  which comprises a low noise amplifier (LNA) and filter  506  and an analog-to-digital converter  508 . A set of counters or registers  510  is also provided to capture and record statistics relating to the received UWB input signal. The analog front end  504  is primarily responsible for converting the received UWB signal into digital form. 
   The digitised UWB signal output from front end  504  is provided to a demodulation block  512  comprising a correlator bank  514  and a detector  516 . The digitised input signal is correlated with a reference signal from a reference signal memory  518  which discriminates against noise and the output of the correlator is then fed to the defector which determines the n (where n is a positive integer) most probable locations and phase values for a received pulse. 
   The output of the demodulation block  512  is provided to a conventional forward error correction (FEC) block  520 . In one implementation of the receiver FEC block  520  comprises a trellis or Viterbi state decoder  522  followed by a (de) interlever  524 , a Reed Solomon decoder  526  and (de) scrambler  528 . In other implementations other codings/decoding schemes such as turbo coding may be employed. 
   The output of FEC block is then passed to a data sychronisation unit  530  comprising a cyclic redundancy check (CRC) block  532  and de-framer  534 . The data sychronisation unit  530  locks onto and tracks framing within the received data separating MAC (Media Access Control) control information from the application data stream(s) providing a data output to a subsequent MAC block (not shown). 
   A control processor  536  comprising a CPU (Central Processing Unit) with program code and data storage memory is used to control the receiver. The primary task of the control processor  536  is to maintain the reference signal that is fed to the correlator to track changes in the received signal due to environmental changes (such as the initial determination of the reference wave form, control over gain in the LNA block  506 , and on-going adjustments in the reference wave form to compensate for external changes in the environment). 
   Referring now to the analog front end  504  in more detail, in a preferred arrangement the LNA block  506  amplifies the signal received from the antenna or cable coupling. The amplifier design contains a fixed frequency passive filter that rejects signals out side of the FCC/ETSC permitted spectral band (3.1-10.6 GHz), as well as rejecting signals from the 5 GHz UNII frequency band. Rejection of such signal areas prevents strong narrow band transmissions from saturating the subsequent A/D converter. It is particularly important to reject signals that are likely to be co-located with a UWB device, such as 802.11, Bluetooth and mobile phone frequencies. 
   The LNA also contains a switchable attenuator that may be used to adjust the signal level fed to the A/D unit. The attenuator may be controlled directly by both the control processor  536  and the reference signal. The purpose of the attenuator is to avoid input saturation at the A/D unit, while maintaining sufficient sensitivity to detect the received pulse waveform. 
   The reference waveform from the Detector unit may also control the attenuation in real time, allowing different gain settings to be applied to different portions of the multipath signals that are received from a single pulse. The A/D converter  508  may take a variety of forms. In a preferred embodiment the A/D converter  508  is logically configured as a continuous sampler, effectively providing a continuous stream of samples at a suitable rate as determined by the upper frequency of the relevant UWB band and the Nyqust criterion, for example 20 G samples per seconds (20 GHz) for a 10 GHz upper frequency. Physically, however, the A/D module may comprise a bank of samplers, for example 16 to provide 16 samples for each received pulse, successively triggered by a phase tapped clock to provide a snapshot of a portion of a received UWB signals at different phases which may then be used to provide an input to the correlator banks  514  of demodulation block  512 . In this way parallel blocks of signal samples may be provided at a rate of a few hundred megahertz, for example at substantially the pulse repetition frequency (PRF) rate thus effectively reducing the primary digitisation clocks speed to this rate; preferably each block substantially spans the duration of a received UWB pulse. Implementing the sampler as a plurality of parallel sampling circuits operating of a phase tapped reference clock facilitates the fabrication of suitable sample (and hold) devices on conventional silicon processors. 
   Some examples of fast A/D converters are the described in the following documents, which are hereby incorporated by referenced: “A 20 GSamples/s 8-Bit A/D Convertor with a 1 MB memory in 0.18 μCMOS presented by Brian Setterberg of Agilent Technologies, Inc., at the 2003 IEEE International Solid-State Circuit Conference (ISSCC)”; “A Serial-Link Transceiver Based on 8-Gsamples/s A/D and D/A Converters in 0.25 μm CMOS presented by Chih-Kong Ken Yang, Vladimir Stojanovic, Siamak Modjtahedi, Mark A, Horowitz and William F. Ellersick, IEEE Journal of Solid-State Circuits, Vol 36, No 11, November 2001”; published US Patent Applications 2002 0167373 and 2002 0145484. 
   Depending upon the application the A/D converter may either be a single-bit converter or a multi-bit converter, and may either monitor the received voltage level or the power level in the received signal. The A/D converter  508  may comprise a non-continuous sampler where the sampler is run only around the expected time of arrival of a received pulse (or around a desired time slice when hunting for a received pulse) and is substantially inactive at other times. In this way a high sampling rate may effectively be achieved but with advantages such as reduced power consumption. 
   In general, it is desirable to gain as much information about the input signal as possible, favouring a multi-bit voltage sensitive sampling scheme. However, implementation constraints (physical silicon area and power consumption) mean that such a scheme is preferably only used for devices where immunity to noise (including unexpected narrow band interference) is important, for example where operation in close proximity to an 802.11 system is envisaged. In some arrangements sure bit conversion permits an acceptable compromise. 
   Non-continuous sampling can offset some of the disadvantages of such a sampler, but can constrain the range of possible delay modulation values that can be detected, thereby reducing the potential information that can be carried by each pulse. Such a trade-off is often acceptable in systems where there are many co-located independent pulse transmissions, since the risk of ‘collisions’ between pulses from different transmissions is reduced. 
   Single bit sampling is prone to saturation but offers a significant saving in silicon cost and power consumption and is therefore preferable level based A/D converters benefit from accurate control the input signal gain. The AFE  504  therefore preferably contains counters that monitor statistics of the input signal conversion, recording the number of values recorded in each of the sampling levels over some period of time. Software running on the Control Processor periodically reads these values and resets the counters. The software may then use these to determine an optimium setting for the gain/attenuation control applied to the received signal by LNA unit  506 . For such purposes, the software may assume that the received signal is, on average, a gaussian noise signal. 
   Referring now to the demodulator block  512 , this is responsible for extracting a data signal imposed on the pulses by a transmitter. 
   The scheme described here is specifically designed to decode modulation by means of the pulse arrival time or by the pulse phase. It may also be adapted to detect modulation by means of the pulse shape (spectral modulation). 
   The input to the demodulator is a stream of sample data from the AFE  504 ; the output is a stream of decoded data bits. The output data rate is substantially constant fixed by the PRF (Pulse Repetition Frequency) and the number of bits encoded by each pulse. The operating parameters of the demodulator (PRF and bit-encoding) are typically fixed for a given transmitter. However, the demodulator (and other system parameters, such as AFE gain) may be time multiplexed by the MAC processor in order to facilitate near simultaneous reception from multiple transmitters. 
   The demodulator contains units to correlate the received signal against a reference signal (programmed and maintained to track changes in the external signal propagation environment) by control processor  536 . The detailed form of the demodulator is shown in  FIG. 6 . 
   Referring to  FIG. 6 , this shows a simplified block diagram of demodulator  512  of  FIG. 5 ; like elements to those of  FIG. 5  are indicated by like reference numerals. The input from the wireless antenna or wired interface and amplifier/filter unit  506  is implemented in discrete analog circuitry, and the A/D (sampler)  508  and demodulator  512  are implemented in the sampling clock domain which has, in one embodiment, an effective range of 25 GHz, corresponding to an actual clock rate of 250 MHz. The system control logic and output to the forward error correction apparatus also operates at 250 MHz. 
   The correlator  514  comprises a bank of multiply-accumulate units  600  each of which receives an input signal sample (comprising a set of samples of the input signal at successive sampling intervals) and multiplies this by a reference sample (comprising a set of samples of a reference waveform at successive sampling intervals) provided by reference waveform synthesiser  518 . In the case of single bit A/D sampling the multiplier operation may be implemented using a simple XOR gate. The accumulators average the (correlation) data over a number of pulses, by averaging over (successive) transmitted pulses bearing the same encoded data and/or averaging over multipath components. 
   The reference signal generator or synthesiser  518  provides the reference signal to the multiply-accumulate units  600  under control of a pattern sequencer  602 . The pattern sequencer is controlled by a PSR (Pseudo Random) sequence lock acquisition module  604 , preferably implemented in software as described later. Conceptually the pattern sequencer  602  provides a reference waveform  606  to a plurality of delay units  608  to provide a plurality of successively delayed versions of the reference waveform to multiply-accumulate units  600 . However although illustrated as a pipeline system with multiply-accumulated delay taps equivalent to a sample period to reduce the effective clock speed the reference waveform is preferably applied in parallel to the multiply-accumulate units  600  as described later. Such a parallel implementation is possible because the reference waveform is stored in memory and therefore a parallel set of differently delayed reference waveforms may be read out from the memory substantially simultaneously; implementation of the demodulator would be significantly more complex were delay taps conceptually applied to the incoming received UWB signal sample data since without additional complexity this would not be readily available in the form of successively delayed time windows of samples of parallel in samples. 
   The reference signal for the correlator is programmed into the reference signal generator  518  by software running on control processor  536 , which preferably uses a training algorithm to determine the receiver response (that is, amptitude and phase distortion to a transmitted pulse). The control processor  536  also maintains a clock phase locked to the PRF (Pulse Repetition Frequency) of the transmitter from which signals are being received by using the arrival times of detected pulses relative to an internal timing reference (Local Crystal Oscillator). A power control output  610  from the reference waveform generator may also be employed to gate power to the A/D and sampling circuitry  508  to put this circuitry into a reduced power mode in periods where there is no expected received signal. This is particularly advantageous in systems using a multi-bit A/D since these often have a relatively large power consumption. 
   A multiply-accumulate unit  600  provide outputs to a discriminator  612  which determines the sign and peak value (or values if probabilistic outputs are supplied to the following stage of the (absolute) value maximum accumulator output). The discriminator outputs provide an output data signal identifying the position of a received pulse and the pulse phase (that is, normal or inverted). A constellation decoder maps this position/phase data from the discriminator to an n-bit symbol which is then passed to the forward error correction block  520 . 
   The demodulator  512  has a plurality of interfaces to other parts of the receiver system, each of which is preferably via a data synchroniser  616   a, b, c , such as a register or buffer. Thus the multiply-accumulate units  600  provide an output to the control processor  536  for calibration of the receiver front end (and preferably also the transmission channel) and for location processing to facilitate physical location of a UWB receiver according to known techniques. The interface between the constellation decoder  614  and FEC blocks  520  is preferably also implemented via a buffer. The PSR lock acquisition module  502  preferably has a bi-directional interface to a software control function implemented on control processor  536  to provide functions such as physical location of the receiver, delay tracking, and data (de) whitening. 
   Referring next to  FIG. 7  this shows relative timings of transmitted data pulses and multipath components of such pulses as seen by the receiver. As can be seen from  FIG. 7  a typical delay span for a multipath reflection is between 1 and 100 ns whereas a typical interval between successive transmitted data pulses is between 2 and 10 ns. It can therefore be appreciated that a multipath reflection of a one pulse may arrive following a direct, line of sight transmission of the next pulse, or even of the next few pulses. The multipath reflections may also be phase inverted subject to different path distortions from the direct path. 
   In a simple but less preferred arrangement the multiply-accumulate stages  600  of the correlator only integrate multipath energy over the inter-transmit pulse period so that, for example in  FIG. 7 , multipath components arriving outside the 2-10 ns delay range would be ignored. However in general typical multipath delays are greater than the average inter-transmit pulse period, and thus significant energy may be lost with this approach. The problem is exacerbated if pseudo-random timing jitter is applied to the timing of the transmitted pulses to achieve spectral whitening. 
   For these reasons it is therefore preferable to implement two or more correlator banks, that is banks of multiply-accumulate units  600  as shown in  FIG. 6 , parallel to facilitate pipelining of the pulse integrations. Such parallelism implemented by repetition of the correlator logic but in a preferred arrangement this parallelism is achieved by multiplexing the use of a single set of multiply-accumulate chains  600 , for example by keeping track of distinct sets of accumulator values in a static RAM (Random Access Memory) buffer memory. 
     FIG. 8  shows a schematic diagram of a UWB signal employing a preferred modulation scheme for the above described receiver and which may be generated by a transmitter described later with reference to  FIG. 10 . The signal of  FIG. 8  may be employed in a wireless or wired UWB transmission system. 
   The signal  800  comprises a plurality of wavelets or pulses  802  each of which has either a normal or inverted form to encode a single bit of information data to be transmitted;  FIG. 8  shows two normal (rather than inverted) examples of such pulses. As illustrated, according to a preferred such a wavelet or pulse comprises a positive-going portion  802   a  and negative-going portion  802   b ; the order of these two portions may be reversed to invert the pulse, thus facilitating generation of normal and inverted pulses in hardware. The pulses  802  have a nominal pulse repetition frequency, for example of the order of 100 MHz. However an additional one or more information data bits may be modulated onto signal  800  by varying the precise position (timing) of a pulse dependent upon the data to be transmitted. For various reasons bi-phase modulation of a UWB signal has been the preferred modulation of many applications. However by also varying the pulse position more data may be encoded onto the UWB signal thus increasing the available data rate for the options for forward error correction at a given data rate and hence the range of a signal. In practical schemes it is further preferable to dither the pulse position (in time) in a deterministic manner to further whiten the UWB signal spectrum and hence reduce the overall signal profile and/or facilitate staying within regulatory boundaries. Thus in addition to the precise timing of a pulse being dependent upon variable information data to be transmitted the pulse position may also be dependent upon a pseudo random or PN (pseudo noise) signal. Such a pseudo random sequence is preferably deterministic so that although apparently random once the sequence and start point is known it can be reconstructed in a deterministic manner at the receiver to allow this PN modulation to be effectively subtracted from the received signal or compensated for in other ways. 
   Preferably the PN modulation is greater than the information data modulation since having a relatively small range of pulse positions about an expected pulse position (once the effects of PN modulation have been compensated for) simplifies demodulation of position-encoded data. In one preferred arrangement, described below, the positions a pulse can take in response modulation by information data are separated by one (or more generally an integral number) of reference (and input) UWB signal sampling intervals. Thus in some preferred embodiments a pulse  802  may take one of eight or 16 different positions in time (although other numbers of positions may be employed) and correlator  514  correlates the input signal with reference signals at all of these positions substantially in parallel to, in a parallel operation, locate the actual or most likely position of a received pulse. As shown in  FIG. 8  according to a typical scheme the duration of a single doublet is typically between 50 ps and 100 ps and the correlator bank  514  performs parallel correlation operations over a time window  804  of approximately ins, thus identifying the pulse as being in one of around 16 overlapping positions. The skilled person will understand that the above timings, and the number of parallel multiply-accumulate units  600  of correlator  514  may be varied according to the requirements of a particular implementation or application. 
     FIG. 9   a  shows one example of an MAC frame  900  for use with the receiver  500  when receiving a signal of the type shown in  FIG. 8 . This MAC frame is, however, provided merely for illustrative purposes and many other different frame formats may be employed. The example MAC frame  900  begins with a preamble sequence  902  comprising 32 bits of preamble data, for example pseudo random data for training. This is followed by a 4 byte header comprising a pseudo random sequence identifier and a pseudo random sequence seed (for identifying a starting point in a sequence), for example as a pair of 2 byte values. Different pseudo random sequences may be employed by different transmitters to help avoid collisions between transmitted UWB data signals. The header is preferably structured to give the appearance of noise, and may therefore include a whitening function—for example the pseudo random sequence identifier and seed may each be selected so that the header appears essentially random. The header is followed by payload data  906  which may also be whitened of a fixed or variable length, for example 128 bytes. 
     FIG. 9   b  schematically illustrates the positions of pilot tone pulses within a UWB signal  910  also comprising information-carrying pulses (not shown). In one arrangement one in every 100 pulses comprises a pilot tone pulse and, as can be seen from  FIG. 9   b , these pilot tone pulses occur at regularly spaced intervals to provide a low-level pilot tone within the UWB signal regulatory spectral mask. Optionally the positions (in time) of the pilot tone pulses may be modulated to provide timing jitter, allowing more frequent or stronger pilot tone pulses within the spectral mask, although this is not necessary. 
     FIGS. 10   a  and  10   b  illustrate an example of a UWB transmitter  1000  which may be employed to generate the information data modulated UWB signal  800  of  FIG. 8 . The transmitter structure of  FIG. 10  is provided by way of example only and other transmitter structures may also be employed to generate the UWB signal of  FIG. 8 . For simplicity forward error coding arrangements are not explicitly shown in the figure. 
   Referring to  FIG. 10   a  a clock  1002  operating at, for example, 250 MHz provides a clock signal to a chain of delay elements  1004   a - e  each providing a delay of, in this example, 40 ps. The successively delayed versions of the clock signal are provided to each of a plurality of monostable pulse generators  1006 , each of which also receives an enable and control input from a controller  1008 . When enabled by the controller  1008  a monostable  1006  provides an output pulse doublet; the phase (normal or inverted) of the pulse doublet is also controllable by controller  1008 . The outputs from all of the monostable pulse generators  1006  are combined, in this example in summers  1008  and the combined output is provided to a transmit antenna  1010 . The controller  1008  receives a pseudo random sequence input from a pseudo noise generator  1012 , and also receives a data and control input  1014 , for example from a preceding forward error correction block and from a transmitter control processor. The data and control input receives information data to be transmitted by the transmitter and control signals such as a timing control signal to control when the transmitter is to transmit and/or pseudo noise sequence selection and start point control signals. The controller  1008  may comprise a state machine implemented in either software or dedicated hardware or a combination of the two. 
   In operation the controller  1008  controls the timing of transmitted UWB pulses and the phase (normal or inverted) of these pulses by providing appropriate enable and phase control signals to the monostable pulse generators  1006  which are then triggered to provide output pulses at the corresponding time by the phase tapped clock from clock signal generator  1002 . 
   Referring now to  FIG. 10   b  this shows an example of one implementation of a monostable  1006  for the transmitter of  FIG. 10   a . The monostable comprises two pulse generators  1020   a, b , one providing a positive-going pulse, the other providing a negative-going pulse, outputs from these two pulse generators being combined in a summer  1022  to provide a pulse doublet output signal  1024 . Both of pulse generators  1020   a  and  1020   b  are controlled by a common enable line  1026  which when asserted enables the pulse generators to provide an output pulse in response to an input timing reference signal on line  1028 , but which when de-asserted disables the pulse generators from providing their outputs. In addition pulse generator  1020   b  has a delay signal input  1030  which delays the production of its output pulse by two cycles to effectively invert the pulse doublet. Thus according to whether or not the delay input  1030  is asserted a pulse doublet comprising either a positive or negative-going pulse or a negative then positive-going pulse is provided. A UWB transmitter such as a transmitter  1000  of  FIG. 10  may be combined with the UWB receiver of  FIG. 5  to provide a UWB transceiver. In this case it is preferable that the UWB transmitter and receiver portions of the transceiver are synchronised to a common PRF clock to avoid self-collision, that is to avoid jamming reception of transmissions from a remote transmitter by local transmissions. 
   Referring next to  FIG. 11 , this shows details of the receiver  500  of  FIG. 5 , and in particular details of the signal acquisition and locking system, including details of the reference signal capture signal. Like elements of those to  FIGS. 5 and 6  are shown by like reference numerals. Broadly speaking the functions of the PSR lock acquisition module  604  are provided by a phase control processor and the functions of the pattern sequencer  602  of  FIG. 6  are provided by a combination of a reference waveform data table and of a PSR sequence generator. 
   As previously described the analog front end and A/D converter  504  provides a plurality of examples of a received UWB input signal in parallel to correlator  514  and each set of input signal samples is processed by a correlator comprising one of multiply-accumulate units  600  of correlator  514  to correlate the set of received samples in parallel with sets of reference signals representing differently delayed pulses. The sets of samples defining differently delayed versions of a referenced signal pulse are derived from a waveform of a pulse stored in a reference waveform data table  1100 . A reference received pulse is preferably stored in this table as a pulse shaped for each of a set of multi part components of the pulse together with data representing delay intervals between these multipath components, as shown in  FIG. 11   b . However differently delayed versions of a pulse may be provided by accessing a common wave shape data store for the pulse. As shown in  FIG. 11   b  a reference or template waveform for a single received pulse having a plurality of multipath components comprises sample data  102  for a plurality of successive sample points of a multipath component of a pulse followed by delay data  1104  representing an interval between that multipath component of the pulse and the next multipath component. An input  1106  allows reference waveform data to be written into the referenced waveform data table  1100 . Reference waveform data is provided to the correlator  514  from the data table  1100  under control of a PSR sequence generator  1108  in synchronisms with a PRF clock input  1110 . 
   A phase control processor  1112  provides a PRF clock to sequence generator  1108  and reference waveform data to data table  1100 . The phase control processor includes a processor and non-volatile program memory storing program code for pilot tone identification, to provide a software phase locked loop (PLL), for multipath component identification, and for template wave shape retrieval and storage. A clock  1114  provides a clock signal to the phase control processor and receives tracking data from processor  1112  comprising a time advance/retard signal for controlling the phase of the clock and a frequency increase/decrease for controlling the frequency of the clock when the phase needs to be consistently advanced/retarded. The clock  1114  is thus adjustable to track movement of the receiver with respect to the transmitter by means of systematic adjustment in the clock timing (which are generally small compared with the modulation). As described further below clock  1114  acts as a slave to a similar clock in a remote transmitter and thus acts as a link clock; typically it has a frequency in the range 50-200 MHz. 
   The phase control processor  1112  provides a control output to a UWB transmitter  1116 , such as transmitter  1000  in  FIG. 10 , to control the transmitter to provide a UWB signal from a transmit antenna  1118  for use in training receiver. The control processor  1112  also receives a starter frame input signal  1120  from a MAC state machine implemented in either hardware or software. The phase control processor  1112  further receives a set of inputs  1122 , one from each accumulator of correlator  514 , and a further input  1124  from the output of discriminator  612 . 
   Broadly speaking, in operation the phase control processor  1112  programs the reference waveform data table  1100  with an initial, predetermined wave shape and then identifies the UWB signal pilot tone and runs a software phase lock loop to lock onto this tone to provide a time reference. The processor then uses this to identify the wave shape of a received pulse, including its multipath components. Optionally the processor  1112  may apply a Fast Fourier Transform (FFT) filter to remove narrow band interference. Broadly speaking to locate the multipath components of a transmitted pulse the phase control processor  1112  scans a sample window by shifting the phase of the PRF clock with respect to the internal clock from clock generator  1114 , integrating to obtain an average sampled data wave shape. Initially the multipath component with the strongest signal is identified and the shape of this multipath component of the pulse determined from the input data, and then the processor hunts for other multipath components both backwards and forwards from the strongest signal (because the direct line of sight pulse may not be the strongest). As previously described the correlator operates with blocks of eight or 16 samples and these blocks are effectively positional in time with respect to the link clock reference from clock generator  1114 . Preferably the multipath component pulse tracking procedure is repeated at a frequency in the order of kilohertz in order to track variations in the multipath channel and, in embodiments where implemented, to obtain physical location information relating to the receiver&#39;s position. In wired UWB transmission systems the multipath environment may be quasi static in which case a channel characterisation procedure such as that described above may only be applied at switch on or, for example, when the error rate increases above a threshold. 
   In the arrangement shown in  FIG. 11   a  the phase control processor receives sampled input signal data via the correlator  514 . This simplifies the architecture of the receiver, although in other arrangements processor  1112  may receive sampled input signal data directly from analog front end  504 . To obtain sample input data from correlator  514  the input data may be correlated with a delta function such as a spike or impulse written into the wave form data table. 
     FIG. 12   a  shows a flow diagram explaining further the operation of the phase control processor  1112  of  FIG. 11   a . To initial calibrate the receiver front end the control processor, at step S 1200 , instructs transmitter  1116  to local UWB pulses under control of the local clock generator  1114 . These pulses are received at a very high signal level and, moreover, processor  1112  knows when these pulses are transmitted and thus knows at what position in time the received input data is expected to comprise a transmitted pulse (taking account of the delay introduced by the separation between transmit antenna  1118  and receive antenna  502  (typically one or a few centimeters)). 
   At step S 1202  processor  1112  programs wave form data table  1100  with a predetermined template, in particular an impulse, and hunts for the transmitted pulses by controlling the timing of PSR sequence generator  1108 . This is conveniently performed by inhibiting generation of a pseudo random sequence so that the phase of the output of generator  1108  may be varied by using the PSR seed as a phase offset adjust. Once the locally transmitted pulses are identified the wave shape of a pulse as received and digitised by analog front end  504  is read from correlator  514  and written into the referenced wave form data table to serve as an initial reference wave form. This in effect calibrates out phase and gain non-linearities in the receiver front end. Although the locally received signal is strong the wave shape data written into the data table  1100  may optionally comprise an average of a plurality of received pulses. 
   Once this initial calibration has been performed the phase control processor  1112  has the more difficult task of frequency and phase locking onto a signal from a remote transmitter and of tracking this signal. Thus at step S 1206  processor  1112  controls the receiver to hunt for a signal at the pulse repetition frequency of the remote transmitter, that is at the pilot tone of the remote transmitter. The pilot tone frequency may not be known exactly but in preferred arrangements is limited to a small set of possible frequencies such as 50 MHz, 100 MHz, and 250 MHz and thus the receiver can pick each of these frequencies in turn to look for incoming UWB signals. The process of hunting for a signal at PRF is illustrated in  FIG. 12   b . The receiver system first runs a correlation in a set of windows  1210  spaced by intervals at the PRF frequency, averaging the correlation results over a plurality of such windows and, if no significant correlation is found, slips the windows, at the same frequency, to a slightly delayed position  1212  as shown in timeline (ii) to repeat the correlation and averaging procedure until pulses at the PRF are found. Once the PRF frequency has been found, because the correlator  514  provides a plurality of outputs corresponding to a small range of delays either side of a desired time position it is straightforward to track variations in the PRF. The clock generator  1114  (and the equivalent in the transmitter) is preferably crystal controlled and thus relatively stable and varies only slowly compared with the kilohertz PLL tracking frequency. The more difficult task is to locate the remote transmitter PRF in the first place, particularly as a pilot tone pulse is transmitted for of the order of only one in 100 pulses, and because the UWB signal is relatively low level, especially at longer ranges. These difficulties are addressed by averaging over a relatively long period in order to identify the systematic pilot tone impulses which appear at fixed times and distinguish, for example, from other UWB pulses which appear effectively at random times. Depending upon the strength of the UWB signal and upon the range and transmit channel it may take as long as one or a few seconds to lock onto the relevant pilot tone as the correlator windows are slipped, which allows averaging over extremely large number of pulses. 
   Once the phase control processor has locked onto the PRF of the remote receiver the processor can rely on the relative stability of clock generator  1114  and can thus rewrite the referenced wave form data table  1100  with an impulse and average over a plurality of pulses, typically between 100 and 1000 pulses, to determine the reference wave form for the transmit channel, and can then write this into the wave form data table. The number of pulses over which the signal needs to be averaged depends upon the range—one pulse may be enough at one meter and average of  10 4 pulses may be necessary at a range of 30 meters. Once the reference wave form for a first multipath component of a transmitted pulse has been identified the phase control processor  1112  can hunt backwards and forwards from this to identify the next multipath component of the pilot tone, and this can be repeated to determine data for a plurality of multipath components of a transmitted pulse. The number of multipath components for which data is acquired depends upon a trade off between acquisition time and system sensitivity (capturing energy from more multipath components facilitates greater sensitivity but takes longer to acquire). It will be appreciated that once the pulse shapes and delays for multipath components of a pulse have been located in time and samples stored tracking the variations of these over time is relatively straightforward and may be accomplished by periodically averaging over say 100 to 1000 pulses, for example by time multiplexing correlator in a similar way to that described below. 
     FIG. 13  shows details of the reference wave form generation system. The PSR sequence generator  1108  receives control signals from the control processor  1112  comprising a pilot tone to control the timing of the reference wave form generation, and a starter frame signal and a sequence seed to control pseudo random sequence modulation for pulse position dithering, and provides a read timing control output  1302  to a pattern controller  1300 . Referring ahead to  FIG. 15   a , this shows the received multipath components of two successively transmitted pulses  1500  and  1502 , each with a plurality of multipath components  1500   a - c ,  1502   a - c . It can be seen that the multipath components  1500   a, b  of pulse  1500  arrive before the start of pulse  1502  but that the multipath component  1500   c  of pulse  1500  arrives between multipath components  1502   a  and  1502   b  of pulse  1502 . In order to correlate the received signal with a reference wave form corresponding to pulse  1500  (or  1502 ) the reference wave form data table  1100  should preferably be able to provide the appropriate multipath component of the pulses at the appropriate times even when these are interleaved as shown. Although this is not essential it is preferable in order to be able to retrieve energy from more multipath components of a received signal. 
   Referring back now to  FIG. 13   a  pattern generator  1300  provides a plurality of outputs  1304  for providing reference wave forms corresponding to a plurality of transmitted pulses having overlapping multipath components. Thus, for example, if it is desired to process overlapping or interleaved multipath components from two successive transmitted pulses pattern controller  1300  provides two address outputs  1304  for addressing the wave form data table at appropriate times to provide portions of the reference wave form corresponding to the overlapping or interleaved portions of the multipath components. Thus referring again to the example of  FIG. 15   a  pattern controller  1300  provides a first address output for controlling data table  1100  to provide multipath components  1500   a, b, c  and  a  second address output for addressing the table to provide the reference wave shapes for multipath components  1502   a, b, c  at appropriate times. It will be appreciated that the number of address outputs of pattern controller  1300  depends upon the delay span of the number of significant multipath components of a pulse as compared with the inter-transmit pulse spacing. The reference wave form data table  1100  provides an output  1306  which comprises a time ordered combination of the multipath components of successfully transmitted components in the example of  FIG. 15   a  multipath components  1500   a ,  1500   b ,  1502   a ,  1502   c ,  1502   b  and so forth. In a preferred arrangement a single set of outputs provides these multipath components in a time multiplexed fashion for use with correlator  514  also operating in a time sliced or multiplexed configuration. However an alternative arrangement is illustrated in  FIG. 13   b  in which data table  1100  has a plurality of sets of outputs, one for each transmitted pulse the receiver is concurrently able to process, which are combined in a summer  1310  and provided as a combined output for subsequent correlation. 
   Referring in more detail to the parallel data outputs from the reference wave form data table, the data table memory is configured to provide a plurality of blocks of reference signal data in parallel, each block of data being delayed with respect to a previous block of data. A block of data may comprise, for example, eight or 16 sample values of the stored reference wave form, preferably defining a multipath component of a pulse such as a one of components  1500   a, b, c  of  FIG. 15   a . The blocks preferably overlap in time and in one arrangement each block is delayed from the previous block by one sample, 16 blocks defining 16 successfully delayed multipath pulse components being output in parallel. In this example this requires a BUS comprises 256 parallel outputs from reference output data table  100 , but the majority of these outputs may be provided simply by appropriate wiring since 16 blocks each of 16 samples, each delayed by a sample requires only 32 parallel sample value outputs. Each of these sample value outputs, it will be appreciated, may comprise a single or multi-bit value, depending upon whether or single or multi-bit A/D conversion is employed. Depending upon the duration of a multipath component of a pulse such as multipath component  1500   a  of  FIG. 15   a  is stored within the reference wave form data table, a block of reference data may be added with zeros at either or both ends. The use of a reference wave form data table provides important benefits to the receiver system, in particular allowing use of a lower quality receiver analog front end than would otherwise be acceptable as the above described process of self-calibration, storing referenced wave form data table  1100 , can compensate for distortion within the receiver as previously described. 
   In operation the PSR sequence generator  1108  is responsive to the pseudo random sequence employed for transmitting the data to control the read timing from the reference wave form data table to compensate for the pseudo random (but deterministic) time modulation imposed on the variable, information—dependent phase and position modulation. Pattern controller  1300  also provides an end of pattern output signal  1308  for use in resetting the correlator as described further below. 
     FIG. 4  shows details of the configuration of the multiply-accumulate units of correlator  514 . The correlator comprises a plurality, in one configuration  16 , of multiply units  1400  each coupled to a respective accumulator  1402 . Each multiplier unit  1400  receives the same block  1404  of sampled input data, as illustrated comprising 16 successively delayed samples (either one or multi-bit values). Each multiply unit  1400  also receives a block of reference signal samples  1406 , in one configuration comprising 16 successive samples of the reference signal wave form, from data table  1100 , but each of blocks  1406  is successively delayed so that the sampled input data is correlated in parallel by multiplier units  1400  with portions of the referenced signal wave form spanning a range (as illustrated, 16) of successive time slices of the referenced wave form. The effect of this is to slide the sampled input data block or time slice along the referenced wave form until a correlation is found but it is easier in practice to firstly change the referenced wave form delay rather than the sampled received data delay, and secondly to perform a plurality of correlation in parallel rather than employ a single slide window. 
   Each of multiply units  1400  comprises a multiplier to multiply each input data sample with the corresponding reference data sample and provide an output to the corresponding accumulator  1402  so that the accumulator accumulates a correlation value from all (in this case 16) correlation operations in parallel. Each accumulator has an output  1408  coupled to a partial correlation store  1410  for writing an accumulated correlation value into the store. Each accumulator also has an input  1412  from a read output of store  1410  to allow a partial correlation value written into the store to be read back from the store and added to a further correlation value in each respective accumulator. Writing of data into the store and reading of data from the store is controlled by the phase control processor  1112 . The partial correlation store  1410  comprises a plurality of sets of memory locations, each set of memory locations storing a set of partial correlation values, one from each multiply-accumulate module (T1 . . . T16). Storage is provided for as many sets of partial correlation values as is needed to process a desired number of transmitted pulses as overlapping or interleaved multipath components. Thus, for example, two sets of memory locations for partial correlation values are provided for storing partial correlation values where multipath components of two successively transmitted pulses overlap or interleave. 
   Data from each of the plurality of memory locations of a set of partial correlation results is provided on an output  1414  to discriminator module  612 . Discriminator  512  also provides a memory clear output  1416  for clearing or setting to zero a set of partial correlation values, and receives an end of pattern signal  1308  from pattern controller  1300 . Discriminator  612  provides an output  1418  to subsequent forward error correction modules such as a Viterbi decoder. Although reference has been made to store  1410  storing partial correlation, once the correlation of a complete set of multipath components of a received signal pulse is complete the accumulated correlation values from outputs  1418  are written into store  1410  thus providing a set of complete correlation values, that is taking account of all multipath components it has been decided to process, and these complete correlation values are available to the discriminator  612  via BUS  1414 . 
   To illustrate the operation of the correlator  514  of  FIG. 14  it is helpful to refer to  FIG. 15   a . Broadly speaking the procedure is to correlate (accumulate) the first received multipath component  1500   a  and to dump this into store  1410 , and then to correlate the next multipath component  1500   b , also accumulating the previously stored partial correlation for multipath component  1500   a  by reading this from store  1410  adding this to the partial correlation value of multipath component  1500   b , and the total accumulated set of correlation values is then written back into store  1410 . This process is continued until a multipath component of a subsequent pulse is encountered, in this case multipath component  1502   a  of pulse  1500 . The pattern controller  1300  of  FIG. 13  then controls the reference wave form data table  1100  to provide a pulse shape appropriate for correlating with multipath component  1502   a  and following the correlation operation the result of this correlation is dumped into a separate set of memory locations within store  1410 , this set of memory locations being allocated to the second pulse. The correlation operation for multipath components of the received signal continues with the partial correlation results being written into the set of memory locations for either the first or second pulse as appropriate, the pattern generator controlling the wave form data table to generate a reference wave shape for the appropriate multipath component. Thus continuing with the example of  FIG. 15   a  multipath component  1500   c  of the first pulse is next accumulated with the partial correlation value read from store  1410  for the first pulse and dumped back into store  1410 . In this case this is the final processed multipath component pulse of  1500  though the accumulated correlation values in store  1410  for the first pulse can then be taken as complete correlation values and processed by discriminator  612 . The signal indicating that the complete set of multipath components has been correlated is provided by pattern controller  1300  since this controller is able to determine that the final stored multipath component has been processed. However correlation of pulse  1502  continues with multipath component  1502   b  and when the first multipath component of a third pulse (is not shown in  FIG. 1500   a ) received the set of partial correlation values which was previously used for pulse  1500  (and which was cleared by discriminator  612  after the complete correlation values for pulse  1500  were processed) is available for use for accumulating correlation values for this third pulse. 
     FIG. 15   b  shows, diagrammatically, the correlation of a multipath component  1510   a  of a received UWB signal pulse  1510  with a set of referenced pulses  1512   a, b  of which, for clarity, only two are shown. The referenced signal pulses are time shifted to either side of the received multipath component  1510   a  and correlation with each of these referenced signal pulses provides a correlation value as schematically illustrated in graph  1514 . The shape of this curve, and the height and width of its peak may alter depending upon the received signal and referenced signal shape. In  FIG. 15   b  a set of (full) correlation values output from storage  1410  to discriminator  612  on BUS  1414  is diagrammatically illustrated by bar chart  1516  in which each bar  1518  represents an accumulated correlation value for one of the delayed versions of the referenced signal multipath component  1512 . It can be seen that most of the accumulated correlation values are close to a mean level  1520  but one of the accumulated values represented by bar  1522  is significantly different from the others. This represents the most likely pulse position; the bars  1524 ,  1526  to either side of it represents next most probable pulse positions. Bar  1522   a  is significantly greater than the average  1520  which applies a positive correlation (normal pulse) whilst bar  1522   b  has a correlation value which is significantly less (more negative) than the average which implies a negative correlation that is an inverted received signal pulse as compared with the reference. Thus the correlator of  FIG. 14   b  is able to co-determine both the likely position (in time) of a received signal pulse and also the phase (normal or inverted) of the pulse and hence to co-determine information data modulated to both pulse position and pulse phase simultaneously. The use of both position and phase simultaneously to encode information data significantly enhances the information data carrying capacity of the system. 
   In the above described system the correlator is employed for correlating successive multipath components of received signal pulses. However essentially the same arrangement can also be used for accumulating relation values for successively transmitting impulses carrying the same data. In other words a transmitter and/or receiver may employ redundancy, using two or more transmit pulses to carry substantially the same data, at the receiver processing these as though they were merely multipath components of a single pulse. This reduces the effective data rate (halving data rate where two pulses are received instead of one to transmit a given symbol) but potentially increases the range of a transmission system by providing greater energy per transmitted symbol. Such an arrangement may be employed adaptively, reducing the data rate but increasing reliability where transmission conditions are difficult or at the edge of range of a system. The reduction in effective data rate may be partially compensated for by increasing the pulse repetition frequency, providing that operation within the desired regulatory spectral envelope is maintained; the transmit power may also be adaptively controlled to facilitate this. 
   No doubt alternatives will occur to the skilled person. It will be understood that the invention is not limited to the described embodiments and encompasses modifications apparent to those skilled in the art lying within the scope of the claims appended hereto. 
   Without further elaboration, it is believed that one skilled in the art can, using the preceding description, utilize the present invention to its fullest extent. The preceding preferred specific embodiments are, therefore, to be construed as merely illustrative, and not limitative of the remainder of the disclosure in any way whatsoever. 
   In the foregoing and in the examples, all temperatures are set forth uncorrected in degrees Celsius and, all parts and percentages are by weight, unless otherwise indicated. 
   The entire disclosures of all applications, patents and publications, cited herein and of corresponding Great Britain application No. 0316897.8, filed Jul. 18, 2003, and U.S. Provisional Application Ser. No. 60/518,344, filed Nov. 10, 2003, are incorporated by reference herein. 
   The preceding examples can be repeated with similar success by substituting the generically or specifically described reactants and/or operating conditions of this invention for those used in the preceding examples. 
   From the foregoing description, one skilled in the art can easily ascertain the essential characteristics of this invention and, without departing from the spirit and scope thereof, can make various changes and modifications of the invention to adapt it to various usages and conditions.