Abstract:
A voltage-to-current converter circuit is disclosed. In one embodiment, the present invention includes a first metal oxide semiconductor field effect transistor (MOSFET) stage operable in a low to medium power range. The present invention also includes a second MOSFET stage operable in a medium to high power range. An additive circuit is utilized to add the contributions of both the first MOSFET stage and the second MOSFET stage. A subtractive circuit is further used to subtract either the first MOSFET stage or the second MOSFET stage when both the first MOSFET stage and the second MOSFET stage are operating in the medium power range and outputting current in a voltage-to-current converting circuit.

Description:
RELATED U.S. APPLICATION 
   This application claims priority to the co-pending provisional patent application Ser. No. 60/457,797, entitled “A Rail-To-Rail Input Linear Voltage To Current Converter,” with filing date Mar. 25, 2003, and assigned to the assignee of the present application. This application is hereby incorporated by reference. 

   FIELD OF THE INVENTION 
   Embodiments of the present invention relate generally to electronic circuitry and in particular to voltage-to-current converter circuits. 
   BACKGROUND OF THE INVENTION 
   Voltage-to-current converter circuits are commonly used in the design of analog and digital electronic circuits. Many embodiments of voltage-to-current converter circuits exist, and some have previously been patented. In general, a voltage-to-current converter takes a voltage as its input, and the circuit converts this to an output current. 
   One embodiment of a conventional voltage-to-current converter is shown in  FIG. 1 . This conventional circuit comprises three transistors and a resistor. The input voltage is coupled to the gate of a first transistor (M 1 ). The drain of the first transistor is coupled to a resistor to ground. The source of the first transistor is coupled to the gates of a second (M 2 ) and third (M 3 ) transistor. The drain of the second transistor is coupled to the source of the first transistor. The sources of the second and third transistors are coupled to power. The drain of the third transistor is coupled to the output of this circuit. 
   The conventional circuit generates an output current by biasing the voltage across resistor R to approximately one NMOSFET (M 1 ) threshold voltage less than the input voltage. By using the M 1  NMOS input, the input impedance (the impedance of the gate) is large, and while M 1  remains in the saturated state, the resistor voltage can remain independent of the state of the M 2  PMOS, which generates a V GS  (gate to source voltage) to mirror the resistor current to M 3  and produce the output current at I OUT . The current generated by the converter circuit  100  is shown in Eq. 1 and the input voltage range of the conventional voltage-to-current converter is shown in Eq. 2 and Eq. 3. 
   
     
       
         
           
             
               
                 
                   I 
                   OUT 
                 
                 = 
                 
                   
                     
                       
                         V 
                         IN 
                       
                       - 
                       
                         V 
                         THN 
                       
                       - 
                       
                         
                           
                             2 
                             * 
                             
                               I 
                               OUT 
                             
                           
                           
                             B 
                             1 
                           
                         
                       
                     
                     R 
                   
                   ≈ 
                   
                     
                       
                         V 
                         IN 
                       
                       - 
                       
                         V 
                         THN 
                       
                     
                     R 
                   
                 
               
             
             
               
                 Eq 
                 . 
                 
                     
                 
                 ⁢ 
                 1 
               
             
           
           
             
               
                 
                   V 
                   
                     IN 
                     - 
                     MINIMUM 
                   
                 
                 = 
                 
                   V 
                   THN 
                 
               
             
             
               
                 Eq 
                 . 
                 
                     
                 
                 ⁢ 
                 2 
               
             
           
           
             
               
                 
                   V 
                   
                     IN 
                     - 
                     MAXIMUM 
                   
                 
                 = 
                 
                   
                     V 
                     PWR 
                   
                   + 
                   
                     V 
                     THN 
                   
                   - 
                   
                     V 
                     THP 
                   
                   - 
                   
                     V 
                     DSAT 
                   
                 
               
             
             
               
                 Eq 
                 . 
                 
                     
                 
                 ⁢ 
                 3 
               
             
           
         
       
     
   
   A disadvantage of the conventional technology is the limited input voltage range, which, from Eq. 2 and Eq. 3, is from approximately V THN  (threshold voltage of a N-type MOSFET) to V PWR  (power supply voltage). When working with low supply voltage, V THN  (approximately 0.7V) can be a significant portion of V PWR , leaving a very narrow input voltage range for the circuit. If the voltage-to-current converter is utilized as a linear tuning element, the narrowed input range forces a greater required change in output current for a given input voltage change, or slope, in order to cover the same output current range. Increasing the I OUT  (output current)/V IN  (input voltage) slope can have a negative impact on noise and noise sensitivity in the circuit and the systems of which it is part. For example, this voltage-to-current converter can be used as the front end of a voltage controlled oscillator in a PLL (phase lock loop). With a narrow input voltage range, the frequency versus voltage slope (or K VCO , the gain of the voltage controlled oscillator) of the oscillator must be large compared to the K VCO  using a wider input voltage range, in order to reach the required range of output frequencies. This, however, causes greater sensitivity to noise on the PLL loop filter resulting in higher PLL output jitter. 
   SUMMARY OF INVENTION 
   Thus, a need exists for a voltage-to-current converting circuit that is operational in a low voltage application. A further need exists for a voltage-to-current converter circuit with a wide input voltage range. Embodiments of the present invention provide these advantages. 
   A voltage-to-current converter circuit is disclosed. In one embodiment, the present invention includes a first metal oxide semiconductor field effect transistor (MOSFET) stage operable in a low to medium voltage range. The present invention also includes a second MOSFET stage operable in a medium to high voltage range. An additive circuit is utilized to add the contributions of both the first MOSFET stage and the second MOSFET stage for the output current. A subtractive circuit is further used to subtract either the first MOSFET stage or the second MOSFET stage when both the first MOSFET stage and the second MOSFET stage are operating in a portion of the medium voltage range and outputting current in a voltage-to-current converting circuit. 
   In this embodiment, the low voltage input stage is responsive to input voltages down to zero volts and upwards to V PWR −V TH . The high voltage input stage is responsive to input voltages of V TH  to V PWR . The combination of the stages provides a converter circuit that is responsive to input voltages from zero to V PWR . 
   The resultant output current of the voltage-to-current converter circuit is linear versus input voltage over a wide range of voltage input values. Features of the voltage-to-current circuit include linear compensation achieved by subtracting the changing current from one portion of the circuit to the other, for example, the first and second MOSFET stages. This achieves linearity over the entire input voltage range, bridging from one voltage-to-current converter half operational input range to the other as discussed above. This is referred to as ‘rail to rail’ operation. 
   Another embodiment of the present invention provides a voltage to current converter circuit having a positive channel metal-oxide semiconductor (PMOS) stage operable in a low to medium power range. In addition, the circuit includes a negative channel metal-oxide semiconductor (NMOS) stage operable in a medium to high power range. An additive circuit is utilized to add contributions of both the NMOS stage and the PMOS stage from the low power range to the high power range to provide a rail-to-rail linear voltage-to-current converter. A subtractive circuit is also used to subtract either the NMOS stage or the PMOS stage when both the NMOS stage and the PMOS stage are operating in the medium power range in a voltage-to-current converting circuit. 
   In its various embodiments, the present invention provides a voltage-to-current converting circuit that is operational in a low voltage application. The present invention also provides a voltage-to-current converter circuit with a wide input voltage range. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The accompanying drawings, which are incorporated in and form a part of this specification, illustrate embodiments of the invention and, together with the description, serve to explain the principles of the invention. 
     Conventional Art  FIG. 1  shows an embodiment of a conventional voltage-to-current converter. 
       FIG. 2  shows a circuit diagram of an exemplary voltage-to-current converter circuit according to one embodiment of the present invention. 
       FIG. 3  shows a circuit diagram of an exemplary implementation of a voltage-to-current converter circuit in accordance with one embodiment of the present invention. 
       FIG. 4  shows a graph of simulation results showing I OUT  versus V IN  for example low V PWR  values of an exemplary voltage-to-current converter circuit according to one embodiment of the present invention. 
   

   DETAILED DESCRIPTION 
   Reference will now be made in detail to embodiments of the invention, examples of which are illustrated in the accompanying drawings. While the invention will be described in conjunction with the preferred embodiments, it will be understood that they are not intended to limit the invention to these embodiments. On the contrary, the invention is intended to cover alternatives, modifications and equivalents, which may be included within the spirit and scope of the invention as defined by the appended claims. Furthermore, in the following detailed description of the present invention, numerous specific details are set forth in order to provide a thorough understanding of the present invention. However, the present invention may be practiced without these specific details. In other instances, well-known methods, procedures, components, and circuits have not been described in detail as not to unnecessarily obscure aspects of the present invention. 
   With reference now to  FIG. 2 , a circuit diagram of an exemplary voltage-to-current converter circuit  200  is shown in accordance with one embodiment of the present invention. This exemplary circuit comprises a first  202 , second  204 , and third  206  current source, and a first (R p ) and second (R N ) resistor. The circuit further comprises a first (M 1 )  240 , second (M 2 )  242 , third (M 3 )  244 , fourth (M 4 )  248 , fifth (M 5 )  250 , sixth (M 6 )  252 , seventh (M 7 )  254 , eighth (M 8 )  246 , and ninth (M 9 )  256  transistor. An input voltage  215  is coupled to the gates of the first  240  and fifth  250  transistors. The drain of the first transistor  240  is coupled to the first current source  202  to ground and draws three times Ic, and is coupled to the drain of the second transistor  242 . The gate of the second transistor  242  is coupled to a bias voltage ‘ncbias’  218 . The source of the first transistor  240  is coupled to a second current source  204 . The second current source  204  is coupled to power (V PWR ), sourcing twice Ic. The source of the second transistor  242  is coupled to the gate and drain of the third transistor  244 , and the gate of the eighth transistor  246 . The source of the third transistor  244  is coupled to power. The source of the eighth transistor  246  is coupled to power. The drain of the eighth transistor  246  is coupled to the source of the ninth transistor  256 . The current that flows between the drain of the eighth transistor  246  and the source of ninth transistor  256  is I OUT    260 , where I OUT    260  is the output of the voltage to current converter circuit  200 . 
   The source of the fourth transistor  248  is coupled to the source of the first transistor  240  and to a first resistor (R p ). The first resistor (R p ) is also coupled to ground. The drain of the fourth transistor  248  is coupled to ground. The gate of the fourth transistor  248  is coupled to the gate of the seventh  254  and ninth transistors  256 . The drains of the seventh  254  and ninth transistors  256  are each coupled to ground. The source of the fifth transistor  250  is coupled to the third current source  206 . The third current source  206  is coupled to power (V PWR ), sourcing I C . The source of the fifth transistor  250  is also coupled to the source of the sixth transistor  252 . The drain of the fifth transistor  250  is coupled to a second resistor (R N ) to ground. The gate of the sixth transistor  252  is coupled to a bias voltage ‘pcbias’  230 . The drain of the sixth transistor  252  is coupled to the gate and source of the seventh transistor  254 . 
   In operation, circuit  200  combines the output currents of a first MOSFET stage  210  and a second MOSFET stage  220  to achieve an input voltage range from 0V to V PWR . Neglecting the presence of the body effect in the first MOSFET stage  210  and the second MOSFET stage  220 , the output current versus input voltage slopes for both halves of the circuit are matched by using equivalent load resistors in each portion. Where the input voltage ranges of both voltage-to-current halves overlap, some circuit compensation is applied to prevent the output current being driven by both, doubling the output current slope over their overlapping input voltage range. In one embodiment, the first MOSFET stage  210  is a PMOS stage  210  and the second MOSFET stage  220  is an NMOS stage  220  of the voltage-to-current converter. 
   In one embodiment, the circuit is compensated by subtracting the output contribution from the second MOSFET stage  220  of the voltage-to-current converter portion of the circuit from the first MOSFET stage  210  portion of the circuit when both stages are active. Therefore, while the first MOSFET stage  210  portion continues to operate, it will remove the output current contributions of the second MOSFET stage  220  portion, via  248 , effectively leaving the output current dependent only on the first MOSFET stage  210  portion. As the first MOSFET stage  210  portion reaches its maximum input voltage, additional increases in the input voltage no longer cause a proportional change in the first MOSFET stage  210  portion output current. In the embodiment shown in  FIG. 2 , this occurs when the first MOSFET stage  210  portion load resistor (R p ) consumes all the load bias current. Since no more bias current is available to source to MOSFET M 1   240  and all further bias current increases due to the compensation circuit flow to the load resistor (R p ), the output current due to the first MOSFET stage  210  portion of the circuit saturates and becomes entirely dependent on the second MOSFET stage  220  portion. The transition between the two portions is smooth since both use identical load resistors (R p  and R N ) (assuming no body effect), and the first MOSFET stage  210  portion “turns off’, when all available current sourced to its load resistor (R p ), with little device distortion (e.g., MOSFET&#39;s falling out of saturation). 
   With reference still to  FIG. 2 , in one embodiment, the circuit  200  is optimized for operation at very low supply voltage and for generating current from a V GND  (negative supply) referred input. The lower supply voltage capability is achieved by indirectly sensing currents by current subtraction in order to minimize the MOSFET threshold contribution in any V PWR  (positive supply) to V GND  (negative supply) path. The generated currents are referred to V GND  by coupling resistors to V GND  and placing the input voltage  215  (with approximately one MOSFET threshold offset) across them. This particular embodiment is also current limited such that it operates rail-to-rail over a fixed range of supply voltage. 
   The embodiment described in  FIG. 2  also utilizes several current sources that may be implemented with one or more MOSFETs whose offsets can be altered. The currents supplied by the sources allow output current variation in the two voltage-to-current halves with some input voltage V IN    215  operation overlap between the two. On the first MOSFET stage  210  side, the bias current and load resistor (R p ) is sized such that the first MOSFET stage  210  portion reaches its maximum input voltage before the load resistor (R p ) voltage pushes any of the bias current sources out of their high impedance current sourcing voltage range (for a MOSFET, out of saturation). An embodiment of the improved circuits using MOSFETs is shown in  FIG. 3 . 
   With reference still to  FIG. 2 , in one exemplary embodiment, a fixed current from current source  204  is applied to the first MOSFET stage  210 . First MOSFET stage  210  generates additional current by using M 1   240  to set a voltage on resistor R p . Resistor R p  will then steal current from fixed current source  204  and push it out the drain of M 1   240  producing a current change that can be measured on M 3   244 . Therefore, if there is no V IN    215  applied to M 1   240 , the bias at M 3   244  is 3*I C . However, if the V IN    215  applied to M 1   240  is low the current mirroring through M 3   244  is I C  (e.g., a bias of 2*I C  is applied to the 3*I C  of the circuit). Therefore, M 3   244  outputs a current range from I C  to 3*I C  depending on the input voltage VIN  215 . In one embodiment, M 2   242  acts as a cascode to separate voltages on the drain of M 1   240  and provides high impedance between M 1   240  and M 3   244 . The following equations are provided as an example of one embodiment of the present invention. 
   Equations 1–3 illustrates the current through M 3   244  based on V IN  at M 1   240 . 
   
     
       
         
           
             
               
                 
                   I 
                   M1 
                 
                 = 
                 
                   
                     
                       2 
                       * 
                     
                     ⁢ 
                     
                       I 
                       C 
                     
                   
                   - 
                   
                     
                       ( 
                       
                         
                           V 
                           IN 
                         
                         - 
                         
                           V 
                           TH 
                         
                       
                       ) 
                     
                     / 
                     
                       R 
                       P 
                     
                   
                 
               
             
             
               
                 Eq 
                 . 
                 
                     
                 
                 ⁢ 
                 1 
               
             
           
           
             
               
                 
                   
                     I 
                     M3 
                   
                   = 
                   
                     
                       
                         3 
                         * 
                       
                       ⁢ 
                       
                         I 
                         C 
                       
                     
                     - 
                     
                       I 
                       M1 
                     
                   
                 
                 ⁢ 
                 
                   
 
                 
                 ⁢ 
                 
                   Where 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   M3 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   244 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   increases 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   with 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   
                     
                       ( 
                       
                         
                           V 
                           IN 
                         
                         - 
                         
                           V 
                           TH 
                         
                       
                       ) 
                     
                     / 
                     
                       R 
                       P 
                     
                   
                 
               
             
             
               
                 Eq 
                 . 
                 
                     
                 
                 ⁢ 
                 2 
               
             
           
           
             
               
                 
                   I 
                   M3 
                 
                 = 
                 
                   
                     I 
                     C 
                   
                   + 
                   
                     
                       ( 
                       
                         
                           V 
                           IN 
                         
                         - 
                         
                           V 
                           TH 
                         
                       
                       ) 
                     
                     / 
                     
                       R 
                       P 
                     
                   
                 
               
             
             
               
                 Eq 
                 . 
                 
                     
                 
                 ⁢ 
                 3 
               
             
           
         
       
     
   
   In general, the above description and equations show how input voltage (V IN    215 ) affects the current I M3  through M 3   244 . In one embodiment, the V IN  input range of M 1   240  is 0 volts to V MAX =V PWR −V TH . The current through M 3  is mirrored to M 8 . 
   With respect still to  FIG. 2 , and now to the second MOSFET stage  220 , in one embodiment, the operation of the second MOSFET stage  220  is similar to that of first MOSFET stage  210  except it operates on the medium to high voltage input  215 , wherein the first MOSFET stage  210  operates on the low to medium voltage input  215 . Moreover, in one embodiment, the bias voltage ‘pcbias’  230  is used to set a bias on M 6   252  such that M 5   250  drain voltage remains reasonable. Without M 6   252 , the voltage felt on the drain of M 5   250  would be biased based on the diode connected M 7   254  which would significantly reduce the input voltage range of M 5  by taking it out of saturation. By using M 6   252 , the voltage maintained on the drain of M 5   250  can be fairly close to V PWR , keeping M 5  saturated over the entire input voltage range of M 5 . 
   The final addition of the currents (e.g., I O    260 ) from both first MOSFET stage  210  and second MOSFET stage  220  comes from the transistors M 8   246  and M 9   256  (e.g., the additive circuit). This current output (e.g., I O    260 ) may be measured or used either directly or by mirroring the current I O    260  to another circuit that could then measure or utilize the converted current. However, another connection is necessary to resolve the current spike in I O    260  that would occur when both the first MOSFET stage  210  and the second MOSFET stage  220  are operational in the mid-voltage range. That is, as stated herein, both the first MOSFET stage  210  and the second MOSFET stage  220  have overlapping ranges, and if left unadjusted, at the overlap a current spike would occur in the slope of the graph of the input voltage versus the output current (e.g.,  FIG. 4 ). 
   For example, the slope of the input voltage versus the output current is based on the equation slope (e.g., I O    260 )=V IN /R. Thus, the slope is determined by R where R=R N =R p , and it is appreciated that without further circuitry, when both are working the overlap current versus voltage slope would double. In order to resolve the spike in I O    260  when both the first MOSFET stage  210  and the second MOSFET stage  220  are operational a subtractive circuit is used. In one embodiment, the current in M 7   254  (and therefore M 8   256 ) is mirrored to M 4   248  (the regulator) and then subtracted from the first MOSFET stage  210  of the circuit  200 . Therefore, at the point that both the first MOSFET stage  210  and the second MOSFET stage  220  are operational, the current generated by the second MOSFET stage  220  is subtracted from the first MOSFET stage  210  via M 4   248  and it will therefore effect equal change to Mg  246  and M 9   256 . That is, any gains of the second MOSFET stage  220  are removed by the first MOSFET stage  210  via M 4   248  thereby keeping the voltage-to-current slope linear over the input range. 
   The counter affect of the first MOSFET stage  210  versus the second MOSFET stage  220  is maintained until the end of the first MOSFET stage  210 &#39;s operational input voltage range. Then when all source  204  current is diverted to resistor R p , no current flows through MOSFET M 1  and the first MOSFET stage  210  effectively turns off. Since the first MOSFET stage  210  was canceling the current from second MOSFET stage  220 , when the first MOSFET stage  210  shuts off, the second MOSFET stage  220  seamlessly takes over the generation of I O    260 , as the current being removed by M 4   248  is no longer being removed. 
   With reference now to  FIG. 4 , a graph  400  of output current (with added offset for display purposes, e.g., separation of 3 graphed portions) versus input voltage is displayed for the embodiment shown in  FIG. 3 . This output demonstrates the exceptionally linear output current versus input voltage characteristic of the improved circuit over several positive power supply (V PWR ) voltage values. 
   Graph  400  is a linear response of the current output to the voltage input. As is shown in graph  400 , the rail-to-rail voltage-to-current is linear throughout the entire range of the voltage inputs shown. Not only does graph  400  show a linear output current versus input voltage over the entire voltage range of the power supply voltage, it further shows the linear current versus input voltage over the entire range of LOW power including below an NMOSFET threshold voltage (in one example an NMOSFET threshold is 0.7V). Therefore, the present invention operates linearly at voltages below which other voltage-to-current conversion circuits are even operational. 
   The circuit of the present invention may be used, for example, to generate the control current in a current controlled oscillator when an effective voltage controlled oscillator is desired. The improved circuit can be utilized to generate a current proportional (with or without a fixed offset) to an input voltage for any application required. 
   An advantage of the circuit of the present invention is that it provides a much greater linear input range voltage range than other voltage-to-current converters and does so without using an amplifier in a compensated feedback loop. This allows the voltage-to-current conversion to operate much faster and without the complexity of a high gain amplifier or ensuring feedback loop stability. Both of these features make the improved circuit highly desirable in a PLL (phase lock look) or DLL (delay lock loop) design as the front end of a voltage controlled oscillator. 
   In alternative embodiments of the improved circuit, the load resistors utilized in the improved circuit can be implemented using any integrated circuit process resistor types (such as, but not limited to, diffusion, polysilicon, or metal), an active resistance composed of a plurality of MOSFETs, bipolar transistors, and diodes, or with a discrete resistor. These embodiments and others not explicitly specified here are envisioned by this improved embodiment. 
   One embodiment of the improved circuit may be implemented using MOSFETs. In another alternative embodiment, this improved circuit may be implemented by interchanging NMOSFETs and PMOSFETs, and V PWR  for V GND . In yet another embodiment, some or all MOSFETs may be replaced with bipolar transistors. In yet another embodiment the current sources shown in  FIG. 2  may be implemented with a plurality of MOSFETs or bipolar devices, or may be provided externally by the combination of one or more currents. 
   Features of the improved circuit include linear compensation achieved by subtracting the changing current from one stage of the circuit to the other. This achieves linearity over the entire input voltage range, bridging from one voltage-to-current converter half operational input range to the other. This is referred to as ‘rail to rail’ operation. 
   A further feature of improved circuit is that it may replace voltage-to-current converters in existing voltage-controlled oscillators to provide a much wider input voltage range. By having a wider input voltage range, the frequency versus control voltage slope (K VCO ) for the oscillator can be reduced which can reduce jitter in phase locked loops (PLLs) using this oscillator. This improved circuit is valuable in very low supply voltage IC processes to provide a maximum input voltage tuning range. 
   Thus, embodiments of the present invention provide a voltage-to-current converting circuit that is operational with good range in a low voltage application. The present invention also provides a voltage-to-current converter circuit with a wide input voltage range. 
   It should be appreciated that reference throughout this specification to “one embodiment” or “an embodiment” means that a particular feature, structure or characteristic described in connection with the embodiment is included in at least one embodiment of the present invention. Therefore, it is emphasized and should be appreciated that two or more references to “an embodiment” or “one embodiment” or “an alternative embodiment” in various portions of this specification are not necessarily all referring to the same embodiment. Furthermore, the particular features, structures or characteristics may be combined as suitable in one or more embodiments of the invention. 
   Similarly, it should be appreciated that in the foregoing description of exemplary embodiments of the invention, various features of the invention are sometimes grouped together in a single embodiment, figure, or description thereof for streamlining the disclosure aiding in the understanding of one or more of the various inventive aspects. This method of disclosure, however, is not to be interpreted as reflecting an intention that the claimed invention requires more features than are expressly recited in each claim. Rather, as the following claims reflect, inventive aspects lie in less than all features of a single foregoing disclosed embodiment. Thus, the claims following the detailed description are hereby expressly incorporated into this detailed description, with each claim standing on its own as a separate embodiment of this invention.