Abstract:
A system for implementing linearization of a radio frequency (RF) power amplifier (PA) in a base station, as well as various component circuitry for implementing said system. By means of a smart partitioning of the signal processing for predistortion between the analog domain and the digital domain, a more linear relationship between the digital input data and the output RF signal is achieved. Linearization of the PA&#39;s output signal is obtained using a mixed-signal apparatus. The digital baseband signal enters the RF signal source. The RF signal source comprises an in-band predistortion circuit, a micro-controller and digital modulator. The output of the digital modulator is an RF signal that enters the PA module. The PA module is composed of the PA and the RF power amplifier linearizer (RFPAL). The RFPAL comprises an RF predistortion circuit, and RF signal analyzer and a microcontroller. In addition, a backward data interface connects the RF signal source with the RFPAL.

Description:
RELATED APPLICATION 
       [0001]    The present invention is a continuation of U.S. patent application Ser. No. 13/250,635, filed Sep. 30, 2011. The disclosure of the foregoing United States Patent Application is specifically incorporated herein by this reference for all purposes as if fully set forth herein. 
     
    
     FIELD OF THE INVENTION 
       [0002]    This invention relates to power amplification. In particular, this invention relates to linearization of a radio frequency (RF) power amplifier, as well as various component circuitry for implementing said system. 
       BACKGROUND OF THE INVENTION 
       [0003]    A telecommunication system consists of various geographically separated nodes exchanging signals. For example, a cellular telephone system includes towers each housing a base station that transmits and receives RF signals to and from cellular telephone transceivers within the service area of the base station. Signals transmitted over a radio link are attenuated due to distance and such factors as propagation loss and multipath fading. Since the strength of the signal is attenuated during transmission between nodes, signals typically transmitted with significant power, using circuit elements known as power amplifiers (PAs). 
         [0004]    Cellular telephone systems employing 2.5G and 3G use sophisticated, non-constant envelop modulation techniques. Examples of such modulation techniques are wide-band code division multiple access (WCDMA), orthogonal frequency division multiplexing (OFDM), multicarrier Global System for Mobile Communications (GSM), and Enhanced Data Rates for GSM Evolution (EDGE). Since the data is encoded under these modulation schemes by amplitude and phase, to achieve highest signal integrity, the output signal of a base station transmitter must be highly linear over a wide dynamic range. The linearity of the PAs is even more important as the cellular phone systems encode higher data rates as more sophisticated systems (e.g., 4G systems) are deployed. Therefore, an ideal PA is expected to pass an input signal through to the output undistorted, with a user-tunable gain and a negligible or minimum delay, and independent of the output impedance of the input signal source. 
         [0005]    A real PA, however, is not ideal over its entire operating range. The deviation from linear input-output relationship in a real PA may result in unwanted amplitude variations of the output signal and which interferes with signals in other radio channels (e.g., by injecting signals of unwanted harmonics at adjacent radio frequency ranges). A cellular wireless communication systems, for example, has a need for a highly linear PA to provide an output signal that achieves a high adjacent channel leakage ratio (ACLR) and a low error vector magnitude (EVM). 
         [0006]    To suppress unwanted PA nonlinearity, a predistortion circuit is provided to model the PA&#39;s gain and phase characteristics. The predistortion circuit provides a pre-distortion signal, which is then combined with the PA&#39;s input signal at the input of the PA. Correctly modeled, the output signal of the PA from the combined signal is that of an overall system that is more linear, as compared to the same system without the contribution of the predistortion signal. Thus, purposely introduced predistortion into the input signal of the PA corrects non-linearity in the output signal of the PA. One example of such a system is provided in U.S. Pat. No. 7,844,014 entitled “Pre-Distortion Apparatus” that issued on Nov. 30, 2010, incorporated herein by reference. 
         [0007]      FIG. 1  is a functional block diagram for system  100 , which is a linearized mixed-signal power amplifier in a base station. As shown in  FIG. 1 , RF signal source  110  includes digital modulator  105  for modulating baseband data  100  on to a carrier signal (not shown) provided by local oscillator (LO)  108 . Digital modulator  105  may be, for example, a digital-to-RF quadrature modulator. The RF signal from digital modulator  105  is transmitted from RF signal source  110  over forward RF path  120  to PA module  130 A using suitable means (e.g., over a coaxial cable). Many applications require separating RF signal source  110  from PA module  130 . PA module  130  may include RF power amplifier linearizer (RFPAL)  135 , which is coupled to RF path  120  via RF couplers  131  and  132 . RFPAL  135  performs adaptive analog predistortion of the RF signal on RF path  120 , prior to amplification by PA  138 . The output signal of PA  138  is fed back using RF coupler  133  to RFPAL  135  to provide adaptive control. 
         [0008]      FIG. 2  is a functional block diagram of RFPAL  135 . RFPAL  135  has three subsystems, as follows: an RF predistortion block (RFPD)  210 , an RF signal analyzer (RFSA)  220  and a micro-controller  230 . RFPD  210  may be implemented, for example, by a RFPD disclosed in co-pending U.S. patent application Ser. No. 12/939,067 entitled “Analog Signal Processor for Nonlinear Predistortion of Radio-Frequency Signals” that names as inventor Qian Yu and others and was filed on Nov. 3, 2010, incorporated herein by reference. Similarly, RFSA  220  may be implemented, for example, by a RFSA disclosed in co-pending U.S. patent application Ser. No. 12/340,032 entitled “Integrated Signal Analyzer for Adaptive Control of Mixed-Signal Integrated Circuits” that names as inventor Qian Yu and others and was filed on Dec. 19, 2008, incorporated herein by reference. 
         [0009]    RFSA  220  derives from signal RF input  201  and RF feedback  203  a complex-valued error signal that represents the waveform distortion. In addition, RFSA  220  also provides a real-time estimate of the power spectral density (PSD) of the error signal. The PSD and the complex-valued error signal are provided as data signal  222  to micro-controller  230 , which provides coefficient vector  232  to RFPD  210 . Coefficient vector  232  allows RFPD  210  to perform adaptive nonlinear analog signal processing on RF input signal  201 . Because of the limitations inherent in analog circuits, however, it is difficult to realize all the desired improvements by way of signal processing techniques using only analog circuits. What is needed is a system that provides improved linearization of power amplifier in a base station without losing the advantages of the analog signal processing performed by RFPAL  135 . 
       SUMMARY OF THE INVENTION 
       [0010]    According to one embodiment of the present invention, a method partitions signal processing between the analog domain and the digital domain to improve predistortion performance over prior art. 
         [0011]    According to one embodiment of the present invention, a mixed-signal RF power amplification system includes (a) an RF signal source for processing baseband digital data, the RF signal source including (i) an in-band digital predistortion circuit and (ii) a digital-to-RF modulator for modulating the baseband digital data; (b) an RF power amplifier module for amplifying the modulated baseband digital data and producing an RF output signal, the RF power amplifier module including (i) an RF power amplifier and (ii) an RF power amplifier linearizer with an RF predistortion circuit; (c) a signal path for transmitting the modulated data from the RF signal source to the RF power amplifier module; and (d) a data path providing an interface between the RF signal source and the RF power amplifier module. 
         [0012]    In one implementation, the in-band digital predistortion circuit further includes a clipper for limiting the amplitude of the baseband digital data, a first finite impulse response filter for suppressing out-of-band harmonics, and a second finite impulse response filter to compensate for non-flat frequency response of the power amplifier. Furthermore, the in-band digital predistortion circuit may further include a normalized mean squared error (NMSE) computation (NMSE1) of the error between of the base band digital data and an output of the first finite impulse response filter for a measure of crest factor. 
         [0013]    In one embodiment, the RF power amplifier linearizer further includes an RF signal analyzer, which includes a finite impulse response filter for compensating for a delay and a gain tilt of the modulated signal. For example, this finite impulse response filter may have 8 taps controlled by 8 complex-valued coefficients. The RF signal analyzer may also further compute a second NMSE (NMSE2) for measuring the error between output value of the finite impulse response filter and a feed-back value from the RF output signal. The RF signal source may further compute a third NMSE (NMSE3) for measuring the modulation error of the digital-to-RF modulator. The modulation error may be caused, for example, by an imbalance between an in-phase channel and a quadrature channel. 
         [0014]    According to another embodiment of the present invention, a mixed-signal RF power amplification system for splitting predistortion between the digital and analog domains include (a) an RF signal module for processing baseband digital data by in-band predistortion in the digital domain and modulating the baseband digital data to a carrier frequency; and (b) an RF power amplifier module for amplifying the modulated digital data with an RF power amplifier and for processing by predistortion in the analog domain, wherein the predistortion by the RF signal module and the RF power amplifier module linearize the output signal of the RF amplifier. 
         [0015]    The RF signal module may include a polar clipper for modifying the cumulative distribution function of the output signal of the power amplifier, a micro-controller for computing an error vector magnitude (EVM) and varying a threshold value for the polar clipper to achieve a predetermined EVM. 
         [0016]    The present invention is better understood upon consideration of the detailed description below and the accompanying drawings. 
     
    
     
       BRIEF DESCRIPTION OF THE FIGURES 
         [0017]      FIG. 1  is a block diagram showing relevant art regarding linearization of a PA in a base station. 
           [0018]      FIG. 2  is a block diagram showing relevant art regarding an RF PA linearizer, such as that introduced in  FIG. 1 . 
           [0019]      FIG. 3  is a block diagram illustrating linearization of a PA in a base station, in accordance with one embodiment of the present invention. 
           [0020]      FIG. 4  is a block diagram of an example embodiment of a subsystem in the in-band predistortion block described in  FIG. 3 . 
           [0021]      FIG. 5  is a block diagram of an example embodiment of an RF PA linearizer. 
           [0022]      FIG. 6  is a block diagram of an example embodiment of subsystem in the RF PA linearizer described in  FIG. 5 . 
       
    
    
     DETAILED DESCRIPTION 
       [0023]      FIG. 3  is a functional block diagram of system  300  which illustrates linearization of a mixed-signal power amplifier in a base station, in accordance with one embodiment of the present invention. As system  100  shown in  FIG. 3 , RF signal source  110 A includes in-band predistortion (IBPD) block  310 , micro-controller  320  and digital modulator  105 . In addition, backward data interface  330  is provided to connect IBPD block  310  with RFPAL  135 A. 
         [0024]    Backward data interface  330  can be either wired or wireless. For example, if the RF signal source and the PA module are connected by a coaxial cable, then backward data interface  330  may use a low frequency band of the connecting coaxial cable. In one embodiment, backward data interface  330  is a low-speed, bi-directional communication interface for exchanging data between the PA module  130 A and RF signal source  110 A. Backward data interface  330  provides data to micro-controller  320 , which processes the data to provide coefficient vector  340  to IBPD  310 . The output signal of IBPD block  310  in system  300  is provided to digital modulator  105 , which may be implemented, for example, by digital modulator  105  of  FIG. 1 . 
         [0025]    IBPD block  310  improves signal quality in several ways. First, IBPD block  310  performs predistortion in the digital domain on only the in-band signals without creating any significant out-of-band spectral emissions. Linear filtering is one method for in-band predistortion that may be used to compensate the frequency response of a power amplifier in multi-carrier, wideband signal applications. IBPD block  310  may modify the complementary cumulative distribution function (CCDF) of the signal as needed. 
         [0026]    The quality of PA output  350  may be characterized by two parameters: the out-of-band emission level, such as, the adjacent channel leakage ratio (ACLR) and the error-vector magnitude (EVM). ACLR is the ratio of the transmitted power to the power measured in the adjacent channels. EVM measures the difference between the measured waveform and the theoretical modulated waveform. In general, it is much more difficult to meet the requirement on ACLR than for EVM. 
         [0027]    Many commercial wireless modulation systems suffer from high peak-to-average ratios, also known as crest factor (CF). Some important examples of modulation schemes with large CF are WCDMA, OFDM, GSM and EDGE. In the prior art, predistortion methods include crest factor reduction (CFR) that reduce the peak-to-average ratio of the modulated signal. In the prior art, CFR and digital predistortion block operate independently. CFR in the prior art trades off the EVM in a controlled manner for significant improvement in ACLR. However, the inventor of the present invention has discovered that, for many RF power amplifiers, the ACLR performance is more correlated to the shape of the CCDF than to the CF. Prior art CFR algorithms, therefore, often lead to sub-optimal ACRL performance. 
         [0028]    In one embodiment, IBPD block  310  may modify the shape of the CCDF to reduce CF and to improve both ACLR and EVM. In addition, further improvements to PA output  150  may be achieved by predistortion in both digital and analog domains in IBPD  310  and RFPAL  135 A, respectively. The result is the joint adaptation of RF signal source  110 A and PA module  130 A connected via backward interface  330 . In one embodiment, ACLR is minimized under a constrained EVM (e.g., EVM is constrained to less than or equal to a maximum value). This detailed description illustrates the present invention using an implementation with this constraint while not being limited by the constraint. 
         [0029]    According to one embodiment of the present invention,  FIG. 4  shows first subsystem  400  in IBPD block  310 . Subsystem  400  implements the digital portion which shapes CCDF using a clipper threshold r th  as a control variable. As shown in  FIG. 4  baseband data signal  100  is received into polar clipper  410 , which processes baseband data signal  100 . The clipped signal (i.e., baseband data signal  200 ) is successively processed in finite impulse response (FIR) filters  420  and  430 . Assume complex baseband signal r(t)e jφ(t)  is received at polar clipper  410 , which output signal is g[r(t)]e jφ(t) , where g(r) is a limiter function with a clipper threshold of r th , limiter function g(r) being given by Equ. 1(a) and 1(b): 
         [0000]        g ( r )= r ( t ) e   jφ(t)  for | r ( t )|&lt; r   th    Equ. 1(a)
 
         [0000]      and 
         [0000]        g ( r )=| r   th   |e   jφ(t)  for | r ( t )|&lt; r   th    Equ. 1(b).
 
         [0030]    The threshold, r th  is taken from coefficient vector  340 , which is provided by microcontroller  320 . 
         [0031]    FIR  420  may be a 128-tap filter designed for suppressing the out-of-band spectral re-growth due to clipping by polar clipper  410 . FIR  420  is implemented using a fast fourier transform (FFT) and overlap-add technique that is well-known to those of ordinary skill in the art. Ideally, the frequency response is unity at in-band frequencies and zero at the out-of-band frequencies. Transitions at band edges may be implemented by raised-cosine roll-off As shown in  FIG. 4 , the output of FIR  420  is referred to as baseband data signal  300 . In one implementation, the filter coefficients for FIR  420  are provided in coefficient vector  340  by microcontroller  320 . 
         [0032]    FIR  430  may be a  3 -tap filter designed for compensating the non-flat frequency response across the signal bandwidth of PA  138 . The first-order effect of a non-flat frequency response is exhibited in the time domain a gain tilt and phase tilt (i.e., a group delay). In one embodiment, only the gain tilt is compensated in IBPD block  310 , while the second-order and higher-order effects of baseband data signal  200  are ignored. To compensate gain tilt, the impulse response of FIR  430  is a 3-tap FIR having coefficients [−ja,1,ja], where a is a real coefficient proportional to the gain tilt (which is also the inverse of the gain tilt of FIR  510 , to be described below in conjunction with  FIG. 5 ). The output signal of FIR  430  is shown in  FIG. 4  as baseband data signal  400 . As in FIR  420 , the filter coefficients of FIR  430  are taken from coefficient vector  340 , which is provided by microcontroller  320 . 
         [0033]    The IBPD block  310  may also provide a normalized mean square error (NMSE) calculated by NMSE  460  arising from the CF, as shown in  FIG. 4 . NMSE  460  is measured in real-time by comparing baseband data signal  300  with baseband data signal  100 , using difference block  440  and mean square error block  450 . When the clipper threshold, r th , is less than the peak envelope, max[r(t)], NMSE  460  is non-zero and increases monotonically with (max[r(t)]−r th ). 
         [0034]      FIG. 5  is a functional block diagram showing one embodiment of an RFPAL  135 A in accordance with one embodiment of the present invention. As shown in  FIG. 5 , RFSA  200 A includes subsystem  500  which provides a real-time measure of the NMSE of the PA which is further described in reference to  FIG. 6 . Backward data interface  330 , described in reference to  FIG. 3 , connects IBPD block  310  with micro-controller  230 . 
         [0035]    RFPD  210  uses a perturbation-based optimization algorithm to minimize the out-of-band emissions in RF Feedback signal  203 . RFPD  210  can compensate for the nonlinearities in both RF input signal  201  and PA output signal  350 . Because RFPD  210  compensates the nonlinearities in the RF signal, the specifications of nonlinearity-related parameters of digital modulator  105  can be relaxed, thus allowing a chip implementation where RF signal source  110 A is integrated on a single complementary metal-oxide-semiconductor (CMOS) circuit. 
         [0036]      FIG. 6  shows subsystem  500  of RFSA  220 A, in accordance with one embodiment of the present invention. Subsystem  500  in RFSA  220 A uses at least the following parameters to calculate NMSE  550 : (1) y ref , a sampled, digitized representation of RF input signal  201 ; (2) y fb , a sampled, digitized representation of RF feedback signal  203 ; and (3) y gain , a sampled, digitized representation of the complex envelope of RF feedback signal  203 , that is obtained from a quadrature down-conversion followed by analog-to-digital conversion by digital modulator  105 . FIR  510  is an 8-tap filter with complex-valued, software programmable coefficients. The firmware in RFSA  220 A optimizes the coefficients sent to FIR  510  by minimizing NMSE  550 , based on the control variables delay and gain tilt. NMSE  550  is a real-time measure of the NMSE of the PA. In one embodiment, FIR  510  is illustrated herein only using the delay and gain tilt control variables. However, additional control variables can be introduced, so that the FIR can generate high-order effects (e.g. group-delay dispersion) for a wide band signal. In general, FIR  510  replicates the linear memory or linear filtering effect of the PA, so that error signal  520  represents the residual nonlinear distortion of the PA. 
         [0037]    In addition, RFSA  220 A provides a real-time PSD estimate  540 . Because PSD estimate  540  is approximately zero at out-of-band frequencies, RFPAL  135   a  can automatically identify the in-band and out-of-band frequency ranges. Alternatively, frequency-range information may be provided by a base station to RFPAL  135 , thus greatly reduces the complexity of the control firmware in RFPAL  135   a.  IBPD block  310  uses the frequency-range data to determine the frequency response of FIR  420  using filter coefficients taken from coefficient vector  340 . Information regarding the PA-induced distortion—in-band frequency ranges, gain tilt and NMSE  550 —are sent by RFSA  220 A to IBPD  310  via data signal  222 , micro-controller  230  and backward interface  330 . 
         [0038]    The real-time optimization of overall EVM is performed by micro-controller  320  in RF signal source  110 A. The threshold r th  mentioned above with respect to Equ. 1(a) and 1(b) for polar clipper  410  is now described in further detail. The EVM is estimated according to Equ. 2: 
         [0000]        EVM =√{square root over (NMSE460+ NMSE 550+ NMSE 660)}  Equ. 2
 
         [0000]    NMSE  460  is already described above with respect to  FIG. 4 . NMSE  550  is described in reference to  FIG. 5 . NMSE  660  is the real-time modulation error of digital modulator  105  and typically arises from I-channel and Q-channel imbalance. Instead of measuring NMSE  660  in real-time, a conservative estimate, ε, of NMSE  660  can be made, and EVM can be computed according to Equ. 3: 
         [0000]        EVM =√{square root over (NMSE460+ NMSE 550+ε)}  Equ. 3
 
         [0039]    When the clipper threshold, r th , is set, coefficients vector  232  for RFPD  210  is adapted to minimize out-of-band emissions. The corresponding result of the out-of-band emission that is measured by EVM as a function of r th  is designated as F(r th ) where F(r th ) is typically a monotonically increasing function. The corresponding EVM estimated from Equ. 3 is denoted by E(r th ) where E(r th ) is a monotonically decreasing function. F(r th ) is often referred to as a cost function and is used for adaptive control of the RFPD. In one example embodiment, the cost function may be the out-of-band emission of the PSD. In that embodiment, the cost function is completely insensitive to the linear memory effect of the PA. In certain applications, it is difficult to extract the out-of-band emission, and the cost function, F(r th ), can be selected as NMSE  550 . In order to use NMSE  550  as a measure of nonlinear distortion, FIR  510  may be controlled in such a manner as to allow it to replicate the linear memory effect of the PA. 
         [0040]    In one example embodiment, RF signal source  110 A can implement firmware that uses a perturbation-based algorithm according to Equ. 4, where E 0  is the maximum allowed value of EVM: 
         [0000]      Min[ F ( r   th )] for  E ( r   th )≦ E   0    Equ. 4
 
         [0000]    Assuming the monotonic behavior of both F(r th ) and E(r th ), the clipper threshold, r th , is varied, as needed, to make it greater or lesser to satisfy the requirement of Equ. 4. Micro-controller  320  is programmed to meet a preferred EVM by approaching E 0  without exceeding this value and remain fixed, if r th  is not saturated at either end of its range. 
         [0041]    In the embodiments disclosed herein, only one control variable, the clipper threshold, r th , is shown, which provides significant performance improvements. However, other control variables are also possible to achieve optimum shaping of the CCDF. 
         [0042]    The above detailed description is provided to illustrate the specific embodiments of the present invention and is not intended to be limiting. Various modifications and variations within the scope of the present invention are possible. The present invention is set forth in the claims.