Abstract:
A low-noise, linearized double-balanced active mixer circuit is described, including a first input for a local oscillator (LO), a second input for an intermediate frequency (IF) signal, and an output for a resulting product radio frequency (RF) signal. The mixer circuit also includes a feedback transformer circuit for the purpose of improving the intermodulation (IM) performance. The lossless nature of the feedback topology gives the active mixer a lower noise figure (NF) characteristic than is realizable with conventional methods. According to a further embodiment, the mixer circuit includes an additional pair of complementary amplifier transistors for the purpose of further improving the IM performance.

Description:
BACKGROUND OF THE INVENTION 
     Mixers are used in communications circuits for the purpose of generating a modulated carrier for transmission, demodulating a modulated carrier in reception, or converting a signal at some input intermediate frequency (IF) to another output radio frequency (RF) by multiplying two input signals and thereby generating a third. A number of mixer realizations, both passive and active, are known in the art, and double-balanced mixers are known particularly well due to their advantages in the suppression of unwanted spurious signals and the isolation of any one of three ports to the other two, there generally being two inputs and one output. The Gilbert Cell has been the most widely used active mixer circuit for performing the above tasks, usually incorporated within an integrated circuit. It does, however, possess certain limitations in terms of intermodulation (IM) distortion and noise figure (NF) that precludes it&#39;s use in communications systems having demanding performance specifications. The series-shunt feedback mixer delivers a much improved IM performance, but the lossy nature of the feedback topology does not improve the NF performance. 
     Referring to FIG. 1, a schematic diagram of a series-shunt feedback mixer is shown in a form that delivers exceptional overall performance. Here, the mixer is comprised of switching transistors  101 ,  102 ,  104 , and  105 , which are turned on (saturation) and off (cutoff) alternately by a differentially applied local oscillator (LO) signal. By this switching action, a pair of currents generated by amplifying transistors  103  and  106  are divided into four paths, there being two paths for each of two currents. The currents generated by switching transistors  103  and  106  are the result of an input intermediate frequency (IF) signal applied differentially across their respective base connections and the series feedback resistors  110  and  111 . The current source  112  serves to establish the quiescent bias condition of the mixer. The hybrid transformers  115  and  116  combine the four currents from switching transistors  101 ,  102 ,  104 , and  105 , creating a pair of feedback voltages  121  and  122 , as well as an output RF signal  123 . The shunt feedback resistors  107  and  108 , in conjunction with the series feedback resistors  110  and  111  and the amplifying transistors  103  and  106 , form a pair of series-shunt feedback amplifiers which serve to establish the conversion gain and improve the IM performance of the mixer. 
     Those familiar with the art will readily understand that the NF performance of the series-shunt feedback mixer is impaired by the dissipative, or lossy, nature of the feedback topology. This active mixer does offer considerable advantages over the more traditional Gilbert Cell active mixer, especially in terms of signal-handling and performance variations over temperature due to the temperature dependency of the emitter resistance r e , and the tradeoffs that are encountered in receiver and transmitter system design. It has long been desirable that a mixer, either passive or active, be available that has improved IM and temperature performance, and at the same time has an improved NF performance without the expense of added power or complexity. 
     It is the purpose of this invention to further advance the art of feedback mixers by addressing the sources of noise present in the series-shunt feedback mixer, and to therefore provide an active mixer of markedly improved NF performance, while at the same time conserving power consumption and retaining the IM performance and overall sense of simplicity and cost effectiveness of the series-shunt feedback mixer. 
     SUMMARY OF THE INVENTION 
     A lossless feedback double-balanced active mixer circuit with improved intermodulation (IM) and noise figure (NF) performance is described which includes a pair of lossless feedback balanced active mixer circuits, each of which includes a differential pair of switching transistors which divide a controlled current into two paths at a rate determined by an input local oscillator (LO). A hybrid transformer in each lossless feedback balanced mixer, consisting of a centre-tapped primary winding and a secondary winding, combines the two currents to provide a recombined amplified IF signal and an output radio frequency (RF) signal. A third amplifying transistor in each lossless feedback active mixer circuit provides the controlled current, which is determined by an input intermediate frequency (IF) signal. Each lossless feedback active mixer circuit further includes a feedback transformer, comprised of an input winding and a tapped output winding, which compares the input IF signal with the recombined amplified IF signal from the hybrid transformers and applies the difference as a correction to the amplifying transistors, thereby completing a lossless feedback amplifier circuit and in turn improving the IM performance of the mixer circuit. Since the feedback transformer is essentially lossless, it introduces no significant sources of noise to the active mixer circuit, and therefore the NF of the of the lossless feedback active mixer circuit remains unimpaired beyond the NF of the transistors themselves. An additional pair of complementary amplifying transistors may be added to improve the IM performance still further. The connection of the secondary windings of the hybrid transformers of the lossless feedback active mixer circuits effectively cancels the output LO and IF signals and provides an output RF signal. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The invention is described in the schematics of FIGS. 1 to  5 , in which: 
     FIG. 1 schematically illustrates the existing prior art, commonly referred to as a series-shunt feedback double-balanced active mixer; 
     FIG. 2 schematically illustrates the existing prior art commonly referred to as a lossless feedback transistor amplifier; 
     FIG. 3 schematically illustrates the existing prior art commonly referred to as a hybrid transformer; 
     FIG. 4 schematically illustrates an embodiment of a lossless feedback double-balanced active mixer in accordance with the present invention; and 
     FIG. 5 schematically illustrates the addition of a pair of complementary amplifying transistors as a further embodiment of the lossless feedback double-balanced active mixer of FIG.  4 . 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     Designers of radio communication receivers are always concerned with elements of system performance which includes, but is not limited to, intermodulation distortion (IM), noise figure (NF), and power consumption. Historically, the IM performance of communications receivers is improved by methods that invariably require additional power consumption. Amplification stages with feedback methods are widely used as they offer far better IM performance while consuming less power than those not employing feedback. The NF of communications receivers is determined by the NF performance of the first stages of the receiver, which usually have sufficiently low NF and suitable signal gain to overcome the IM and NF performance of the first mixer stage, which is traditionally the primary source of distortion and noise. This invention now presents a mixer circuit which achieves a markedly improved IM and NF performance without excessive power consumption by applying a feedback method widely used in amplifier design which introduces no significant noise sources in addition to those of the active devices themselves. 
     A typical series-shunt feedback double-balanced active mixer is shown in FIG.  1 . Here, transistor  103  and resistor  107  and  110  form a series-shunt feedback amplifier on the left side, while transistor  106  and resistors  108  and  111  form a series-shunt feedback amplifier on the right side. Transistors  101  and  102  form a chopper for the left side and transistors  104  and  105  form a chopper for the right side. Hybrid transformer  115  combines the currents from transistors  101  and  102 , the sum of which appears at the centre tap while the difference appears at the secondary winding. A similar description can be made for the second hybrid transformer  116  on the right side. If both hybrid transformers  115  and  116  have turns ratios of 1:1:1 (K=1), then the signals at the center taps of the hybrid transformers  115  and  116  are, respectively: 
     
       
           V   121   =V   CC   −A×A   V ×Cos ω S   t   (1) 
       
     
     
       
           V   122   =V   CC   +A×A   V ×Cos ω S   t   (2) 
       
     
     where ω S  is the frequency and A is the amplitude of the input Intermediate Frequency (IF) signal and                A   V     =     1   -         R   12         R   11     +     r   e                     (   3   )                                
     where r e  is the incremental emitter resistance of amplifying transistors  103  and  106 . Further, the signal voltages at the collectors of switching transistors  101 ,  102 ,  104 , and  105  are, respectively:                V   117     =       V   CC     +       A   ×     A   V     ×     [         Cos        (       ω   L     -     ω   S       )          t     +       Cos        (       ω   L     +     ω   S       )          t       ]       2               (   4   )                 V   118     =       V   CC     -       A   ×     A   V     ×     [         Cos        (       ω   L     -     ω   S       )          t     +       Cos        (       ω   L     +     ω   S       )          t       ]       2               (   5   )                 V   119     =       V   CC     +       A   ×     A   V     ×     [         Cos        (       ω   L     -     ω   S       )          t     +       Cos        (       ω   L     +     ω   S       )          t       ]       2               (   6   )                 V   120     =       V   CC     -       A   ×     A   V     ×     [         Cos        (       ω   L     -     ω   S       )          t     +       Cos        (       ω   L     +     ω   S       )          t       ]       2               (   7   )                                
     where ω L  is the frequency of the input Local Oscillator (LO) signal. Finally, the output signal is: 
     
       
           V   123   =A×A   V ×[Cos(ω L −ω S ) t +Cos(ω L +ω S ) t]   (8) 
       
     
     Referring now to FIG. 2, a lossless feedback transistor amplifier circuit  200  is shown in its most basic form. Here, a transformer  203 , having a turns ratio of 1:N:M, provides the feedback coupling from the collector to the emitter of transistor  204 . An input signal voltage source  201  provides an input signal  202 , having an amplitude A and a frequency ω S , which causes a current to flow through the input winding of transformer  203  and then into the emitter of transistor  204 . This, in turn, causes a collector current from transistor  204  to flow into the output winding of transformer  203 , thereby creating an output signal voltage  207  to appear across the load resistance  208  (illustrated as a fixed resistance R L  for convenience). Briefly, the collector of transistor  204  sees a load resistance equal to: 
     
       
           R   206 =( N+M )× R   L   (9) 
       
     
     while the input resistance of the lossless feedback amplifier is:                R   202     =         M   +   N   +   1       M   2       ×     R   L               (   10   )                                
     and the power gain in dB is: 
     
       
           G   dB   =N+M+ 1  (11) 
       
     
     Referring to FIG. 3, a circuit  300  is used as an aid in describing the impedances, voltages, and currents of the four ports of a hybrid transformer  301 , which are: 
     
       
           R   304   =K   2   ×R   302   (12) 
       
     
     
       
           R   303   =R   305 =2× R   304   (13) 
       
     
     
       
           I   306   =K ×( I   309   −I   307 )  (14) 
       
     
     
       
           I   308   =I   309   +I   307   (15)  
       
       
         
           
             
               
                 
                   
                     V 
                     306 
                   
                   = 
                   
                     
                       
                         V 
                         309 
                       
                       - 
                       
                         V 
                         307 
                       
                     
                     
                       2 
                       × 
                       K 
                     
                   
                 
               
               
                 
                   ( 
                   16 
                   ) 
                 
               
             
             
               
                 
                   
                     V 
                     308 
                   
                   = 
                   
                     
                       
                         V 
                         309 
                       
                       + 
                       
                         V 
                         307 
                       
                     
                     2 
                   
                 
               
               
                 
                   ( 
                   17 
                   ) 
                 
               
             
           
         
                 
         
             
         
      
     
     Referring now to FIG. 4, a lossless feedback double-balanced active mixer circuit  400  in accordance with the present invention is illustrated. Mixer circuit  400  includes a first pair of switching transistors  401  and  402  and a second pair of switching transistors  404  and  405 . The emitters of switching transistors  401  and  402  are connected in common to the collector of an amplifier transistor  403 , the base of which is grounded. The emitter of the amplifier transistor  403  is connected to one end of an input winding of a lossless feedback transformer  407 . The opposite end of the input winding is connected to receive one of a complementary pair of IF signals thereon. The emitters of switching transistors  404  and  405  are connected in common to the collector of an amplifier transistor  406 , the base of which is grounded. The emitter of the amplifier transistor  406  is connected to one end of an input winding of a lossless feedback transformer  408 . The opposite end of the input winding is connected to receive the other of the complementary pair of IF signals thereon. The bases of switching transistors  401  and  405  are connected together and to receive one of a complementary pair of local oscillator signals thereon. The bases of switching transistors  402  and  404  are connected together and to receive the other of the complementary pair of local oscillator signals thereon. 
     The collectors of switching transistors  401  and  402  are connected to opposite sides of a primary winding of a hybrid transformer  411 . A centre tap of the primary winding is connected to one end of an output winding of lossless feedback transformer  407 . The opposite end of the output winding is connected to a voltage source V CC  and a tap of the output winding is connected through a load resistance  409  (illustrated as a fixed resistance R 41  for convenience) to voltage source V CC . The collectors of switching transistors  404  and  405  are connected to opposite sides of a primary winding of a hybrid transformer  412 . A centre tap of the primary winding is connected to one end of an output winding of lossless feedback transformer  408 . The opposite end of the output winding is connected to a voltage source V CC  and a tap of the output winding is connected through a load resistance  410  (illustrated as a fixed resistance R 41  for convenience) to voltage source V CC . An RF output terminal  421  is connected through a secondary winding of hybrid transformer  411  to ground, through a secondary winding of hybrid transformer  412  to ground, and through resistance designated  422  (illustrated as a fixed resistance R L  for convenience) to ground. 
     The input impedance of mixer circuit  400 , as seen at either of the IF input ports, is determined from EQ. 10 by the value of the resistors  409  and  410 , as well as the turns ratios of the lossless feedback transformers  407  and  408 :                R     i                 n       =         M   +   N   +   1       M   2       ×     R   41               (   18   )                                
     It is necessary that the impedance of the centre tap of hybrid transformers  411  and  412  be matched to the collector load impedance of the lossless feedback transformers  407  and  408 , respectively: 
     
       
           R   419 =( M+N )× R   41 =2× K   2   ×R   L   (19)  
       
       
         
           
             
               
                 
                   
                     R 
                     41 
                   
                   = 
                   
                     
                       2 
                       × 
                       
                         K 
                         2 
                       
                       × 
                       
                         R 
                         L 
                       
                     
                     
                       M 
                       + 
                       N 
                     
                   
                 
               
               
                 
                   ( 
                   20 
                   ) 
                 
               
             
           
         
                 
         
             
         
      
     
     which forces the IF input impedance of both sides of the double-balanced lossless feedback active mixer circuit to be:                R     i                 n       =       2   ×     K   2     ×     R   L     ×     (     M   +   N   +   1     )           M   2     ×     (     M   +   N     )                 (   21   )                                
     These conditions being satisfied, the input currents to the emitters of amplifying transistors  403  and  406  are, respectively:                I   413     =       I   Q     +       A   ×   Cos                   ω   S        t       R     i                 n                   (   22   )                 I   414     =       I   Q     -       A   ×   Cos                   ω   S        t       R     i                 n                   (   23   )                                
     where I Q  is the quiescent bias current for amplifying transistors  403  and  406 , A is the amplitude and ω S  is the frequency of the input IF signal voltage. The current at the collectors of switching transistors  401 ,  402 ,  404 , and  405  are, respectively:                      I   415     =                    I   403     ×       1   -     Cos                   ω   L        t       2       =                 =                      I   Q     ×     (     1   -     Cos                   ω   L        t       )       2     +       A   ×     [       Cos                   ω   S        t     -           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R     i                 n                           (   24   )                       I   416     =                    I   403     ×       1   +     Cos                   ω   L        t       2       =                 =                      I   Q     ×     (     1   +     Cos                   ω   L        t       )       2     +       A   ×     [       Cos                   ω   S        t     +           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R     i                 n                           (   25   )                       I   417     =                    I   406     ×       1   +     Cos                   ω   L        t       2       =                 =                      I   Q     ×     (     1   +     Cos                   ω   L        t       )       2     -       A   ×     [       Cos                   ω   S        t     +           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R     i                 n                           (   26   )                       I   418     =                    I   406     ×       1   -     Cos                   ω   L        t       2       =                 =                      I   Q     ×     (     1   -     Cos                   ω   L        t       )       2     -       A   ×     [       Cos                   ω   S        t     -           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t       2       ]         2   ×     R     i                 n                           (   27   )                                
     The currents at the centre taps of hybrid transformers  411  and  412  are, respectively:                I   419     =         I   415     +     I   416       =       I   Q     +       A   ×   Cos                   ω   S        t       R     i                 n                     (   28   )                 I   420     =         I   417     +     I   418       =       I   Q     -       A   ×   Cos                   ω   S        t       R     i                 n                     (   29   )                                
     and the output current is:                      I   421     =                    K   ×     (       I   415     -     I   416       )       -     K   ×     (       I   417     -     I   418       )         =                 =                2   ×   A   ×     K   2     ×           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t         R     i                 n                         (   30   )                                
     which makes the output voltage equal to:                V   421     =     2   ×   A   ×     K   2     ×     R   L     ×           Cos        (       ω   S     -     ω   L       )          t     +       Cos        (       ω   S     +     ω   L       )          t         R     i                 n                   (   31   )                                
     Referring now to FIG. 5, a further embodiment of a double-balanced lossless feedback active mixer circuit  500  in accordance with the present invention is shown. Mixer circuit  500  includes a first pair of switching transistors  501  and  502  and a second pair of switching transistors  505  and  506 . The emitters of switching transistors  501  and  502  are connected in common to the common connected collectors of a pair of complementary amplifier transistor  503  and  504 , the bases of which are grounded and the common connected emitters of which are connected to one end of an input winding of a lossless feedback transformer  509 . The opposite end of the input winding is connected to receive one of a complementary pair of IF signals thereon. The emitters of switching transistors  505  and  506  are connected in common to the common connected collectors of a pair of complementary amplifier transistor  507  and  508 , the bases of which are grounded and the common connected emitters of which are connected to one end of an input winding of a lossless feedback transformer  510 . The opposite end of the input winding is connected to receive the other of the complementary pair of IF signals thereon. The bases of switching transistors  501  and  506  are connected together and to receive one of a complementary pair of local oscillator signals thereon. The bases of switching transistors  502  and  505  are connected together and to receive the other of the complementary pair of local oscillator signals thereon. 
     The collectors of switching transistors  501  and  502  are connected t o opposite sides of a primary winding of a hybrid transformer  513 . A centre tap of the primary winding is connected to one end of a tapped output winding of lossless feedback transformer  509 , which has turns ratios of M and N with respect to the input winding. The opposite end of the output winding is connected to a voltage source V CC  and a tap of the output winding is connected through a resistance  511  (illustrated as a fixed resistance R 51  for convenience) to voltage source V CC . The collectors of switching transistors  505  and  506  are connected to opposite sides of a primary winding of a hybrid transformer  514 . A centre tap of the primary winding is connected to one end of a tapped output winding of lossless feedback transformer  510 , which has turns ratios of M and N with respect to the input winding. The opposite end of the output winding is connected to a voltage source V CC  and a tap of the output winding is connected through a resistance  512  (illustrated as a fixed resistance R 51  for convenience) to voltage source V CC . An RF output terminal  516  is connected through a secondary winding of hybrid transformer  513  to ground, through a secondary winding of hybrid transformer  514  to ground, and through a load resistance  515  (illustrated as a fixed resistance R 41  for convenience) to ground. The various gain and impedance relationships for mixer circuit  500  are identical to those established earlier for mixer circuit  400 . 
     Although detailed embodiments of the invention have been described, it should be appreciated that numerous modifications, variations, and adaptations may be made without departing from the scope of the invention as described in the claims. For example, those familiar with the art will recognize that the bipolar transistors shown in the embodiments may be alternatively replaced with field effect transistors. Also, the single-transformer lossless feedback topology shown in the embodiments may be alternatively replaced with other forms of lossless feedback that are known to the art. 
     Further, while the terminals of the bipolar transistors described in the various embodiments are referred to as the emitter, base and collector, it will be understood that these terminals will be the source, gate and drain when the transistors utilized are field effect transistors or other similar types and may be referred to as input, control and output terminals, respectively, however the titles of the various components and terminals are only intended to enhance the understanding of the disclosure and are not intended to in any way limit the type of component utilized. In addition, it should be understood that the terms “lossless feedback transformer” and “hybrid transformer” used throughout this disclosure refer to general types of transformers and should not be limited in any way to specific types of transformers.