Abstract:
There is disclosed a phase locked loop comprising: a phase frequency detector for receiving as a first input a reference signal and for generating a control signal; a voltage controlled oscillator for receiving the control signal and for generating a signal defining an output frequency, a feedback path connecting the output signal to a second input of the phase frequency detector; and a digital accumulator for generating the reference signal under the control of an accumulator reference clock.

Description:
FIELD OF THE INVENTION 
   The invention relates particularly, but not exclusively, to an accumulator based, phase locked loop system, such as is typically used in a telecommunications system. However, the invention could equally be applied in other applications relying on a frequency synthesiser or a phase modulated signal source. 
   BACKGROUND OF INVENTION 
   Frequency synthesisers are an integral part of any modern communications system, especially any coherent system dependent upon a steady phase difference between each element in the communication network. 
   Advancing technology has made possible the use of adaptive beam steering using multiple antennas to provide a focused beam between transmitter and receiver systems, allowing the dual benefits of increased cell capacity with increased rejection of interfering signals. The ability to adjust the output phase of the frequency under digital base-band control is particularly useful as it can reduce the component count used in a transmitter system. Similarly a system which can accurately provide a phase modulated radio frequency signal offers the potential for higher levels of component integration. 
   Reference is made to  FIG. 1 , which shows a known synthesiser generally designated by reference numeral  10 . 
   The synthesiser  10  comprises a phase detector  2 , a low-pass filter  4 , voltage controlled oscillator (VCO)  6 , a divider  8 , and an optional integer “R” divider  12 . A signal having a required output frequency is generated at the output of the VOC  6 . 
   Essentially there are two variants to this type of synthesiser, employing either fixed integer dividers or modulated dividers, by a suitably adjusted data stream, which constantly adjusts the integer divider value in the divider  8 . Modulated dividers are also known as fractional-N dividers. The parasitic modulation of the divider output signal is itself modulated by this modulated data stream to ensure the remainder of the phase locked loop can remove this parasitic modulation, whilst preserving the advantages it offers. 
   Both types of phase locked loops use a phase frequency detector  2 , which is commonly a digital element, to compare a fed back voltage controlled oscillator output with an incoming reference signal, to which the system is phase locked. The output of the phase frequency detector  2  can be a series of either current or voltage pulses, which are filtered by the loop filter  4  to give a small error voltage. This small error voltage complements the voltage pedestal at the output of the loop filter, ensuring the VCO remains phase locked. The error voltage is the correction voltage supplied to the VCO, to suppress the excess phase noise of this device to levels determined by the phase locked loop dynamic characteristics. The VCO is the device controlled by this negative feed back closed loop system. To provide a frequency translation back to the phase frequency detector  2 , for comparison with the reference signal, the series divider circuit is used. If fractional-n dividers are used, as shown in  FIG. 1 , their mean division value can be adjusted to give an effective multiplication of the reference signal, hence allowing the phase locked system to change to different frequencies with respect to the reference signal. 
   Digital dividers have the net effect of raising the phase noise of the system because their dividing action in the feed back path of the system translates to a multiplication of PFD related noise in the through transfer characteristic of the phase locked loop. 
   Arrangements using digital dividers in their feed-back path offer limited noise performance preventing their simple implementation in new, more demanding, communication system applications. 
   Single loop fractional-N techniques, described earlier, have been adapted to improve on divider limitations by raising the sampling frequency seen at the digital phase frequency detector input to reduce these division values. However, a point is reached where the sampling frequency approaches half the synthesiser&#39;s output frequency (for a minimum division value of 2). Beyond this point only another 3 dB of improvement might be possible, although this still does not guarantee that the resulting in-band phase noise becomes acceptable. 
   Also known is the use of phase locked loops employing a mixer as a frequency translation element, in order to improve the in-band phase noise using a combination of analogue phase locked loops and direct digital synthesisers. In an alternative PLL arrangement, a mixer arrangement is used in place of the divider  8 . When mixers are used, additional signal sources are required to provide this frequency translation. Mixers do not raise the in-band phase noise levels, because their action is to subtract two signals in the feed back path, giving no change in phase at the mixer output, and hence no adverse effect to a system which tracks only phase. A phase locked loop using a mixer has a minimal multiplication of any spurious energy injected into the reference input. 
   The direct digital synthesisers provide the necessary frequency interpolation required for attaining the specified frequency steps at the phase locked loop output. Reference is made to the Qualcom application note AN2334-4, (1990) and U.S. Pat. Nos. 4,965,533, and 5,184,093 on the subject. 
   Alternatively the direct digital synthesiser is applied to the phase frequency detector input, with a consequent spurious degradation seen at the phase locked loop output. 
   A phase locked loop possesses a typical transfer characteristic of a band-pass system. This band-pass is filter characteristic is centred about the output VCO&#39;s signal, which at high frequencies (given the low loop bandwidths of the phase locked loop) represents a very high “Q” factor that cannot be achieved any other way. 
   The alternative to analogue phase locked loops, as described herein above, are direct digital synthesisers. Direct digital synthesisers are not phase locked systems, as they do not possess a feedback path between their output and inputs. They are capable of open loop operation because their all-digital nature guarantees repeatable outputs under all conditions. They do not suffer from the vagrancies of analogue systems. The basic concept of direct digital synthesisers remains unchanged from the original paper presenting the idea given in 1971. As shown in  FIG. 2 , a direct digital synthesiser  20  is built up of three components; a digital (phase) accumulator  14 , a sine (or cosine) look-up table  16 , and a digital to analogue converter (DAC)  18 . External to the direct digital synthesiser  20  is an analogue low pass or band pass filter  22 . A reference clock  24  clocks the digital accumulator  14  and the DAC  18 . 
   The purpose of the digital accumulator  14  is to digitally integrate the digital input word provided on an input thereto, resulting in a ramp output at the required frequency. This defines the digital input word as a phase value equivalent to the phase difference over one accumulator clock period to give the required output frequency. Every time the accumulator overflows the “carry out” bit is ignored and the accumulator output re-starts it&#39;s integration sequence, giving an output word pattern resembling a ramp. The length of the digital accumulator  14  determines the phase resolution available for each accumulator clock cycle and hence the accuracy of the output frequency. Using this concept of phase increments, the required digital input word for a given output frequency can be calculated using the expression: 
   
     
       
         
           
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   In a practical system the length of the accumulator data word exceeds the resolution of the following sine look up table, therefore only the “P” most significant bits are fed into the sine look up table. The value of “P” depends upon the combined width of the sine look up table and any compression circuitry used to mirror and invert the output of the sine look-up table output. It is the function of the sine look-up table  16  to convert the truncated accumulator equivalent phase value to a digital equivalent amplitude value, using a sine or cosine transfer characteristic. This digital amplitude is converted into an analogue signal level using a digital to analogue converter  18  clocked at the same frequency as the digital accumulator  14 . In some direct digital synthesiser designs additional pipelining circuitry may be added to overcome circuit settling times allowing higher frequencies of operation. There is no effect on the quality of the output signal, with such pipelining only a slight phase delay is incurred between a change in digital input to analogue output. 
   Direct digital synthesisers are comprised of all digital elements making them suitable for integration into a chip. However one major performance limitation is the digital-to-analogue converter at the output. This digital-to-analogue converter generates problems; reducing the spurious free dynamic range and raising the noise floor. To minimise these effects caused by sampling and aliasing during the digital-to-analogue converter operation, the passive reconstruction filter  22  is normally used to “clean-up” the signal before it is used in the remainder of the system it is employed to drive. 
   Direct digital synthesisers cannot operate at the required local oscillator frequencies of contemporary mobile communication systems with the necessary noise and spurious performance. Therefore in current known solutions direct digital synthesisers are combined with analogue or digital phase locked loop techniques, to perform the necessary up-conversion of their lower frequency signals. 
   As described hereinabove, any phase locked loop employing a digital divider in its feedback path possesses gain. Therefore any small direct digital synthesiser related spurious products would be subject to this gain, usually raising their level to unacceptable levels. Analogue loops using only a mixer in the feedback path have no such gain, giving a unity translation of input DDS spurious levels. 
   It is an aim of the present invention to provide a solution which overcomes the above-stated problems. 
   SUMMARY OF THE INVENTION 
   In accordance with the present invention, there is provided a phase locked loop comprising: a phase frequency detector for receiving as a first input a reference signal and for generating a control signal; a voltage controlled oscillator for receiving the control signal and for generating a signal defining an output frequency, a feedback path connecting the output signal to a second input of the phase frequency detector; and a digital accumulator for generating the reference signal under the control of an accumulator reference clock. 
   Preferably there is further provided a summer having a first input connected to the output of the phase frequency detector and an output connected to the input of the voltage controlled oscillator, and a digital to analogue converter having a first input connected to the output of the digital accumulator, a second input of the summer being connected to the output of the digital to analogue converter. 
   Preferably the digital to analogue converter is clocked by a clock signal derived from dividing the accumulator reference clock. 
   The digital to analogue converter may be connected to the digital accumulator via a latch. The latch may be clocked by the clock signal derived from the accumulator reference signal. 
   The accumulator reference clock may be divided on input to the accumulator. The digital to analogue converter may be connected to the digital accumulator via a look-up-table. The look-up-table may be one of either a sine look-up-table or a cosine look-up-table. 
   The phase detector circuit may receive a further reference signal from the digital accumulator. 
   The reference signal may be provided by the most significant bit of the digital accumulator, and the further reference signal is provided by at least one further bit of the digital accumulator. 
   The most significant bit may be provided to the phase detector via a divider, and the at least one further bit is provided to the phase detector by a latch. The latch may be clocked by a divided version of the accumulator reference clock. 
   The feedback path may be provided by a divider. 
   The accumulator reference clock may be provided by a reference clock generator. 
   The feedback path may be provided by a mixer. The mixer may receive as a first input the signal defining the output frequency and as a second input the accumulator reference clock. The accumulator reference clock may be generated by a further phase locked loop. The further phase locked loop may generate the accumulator clock as its output signal and receives a reference clock signal as the input reference signal. The digital accumulator may receive the accumulator reference clock, and the mixer receives a further accumulator reference clock. The accumulator reference clock and the further accumulator reference clock may be provided by first and second further phase locked loops. 
   The further phase locked loop circuit may comprise a reference divider for receiving a reference clock signal, a phase detector connected to receive the output of the reference divider, and a voltage controlled oscillator connected to receive the output of the phase detector, the output of the voltage controlled oscillator forming the second input to the mixer and the clock input for the digital accumulator, there further being a feedback path from the output of the voltage controlled oscillator to the phase detector of the reference generating circuit. 
   The feedback path may comprise a divider. 
   There may further be provided a low pass filter at the input to the voltage-controlled oscillator of the further phase locked loop. 
   The digital accumulator may receive as an input a digital frequency input word. 
   There may further be provided a low pass filter at the input of the voltage-controlled oscillator. 
   There may further be provided an IF filter at the output of the mixer. 
   A mobile communication system may include a phase locked loop as described. A mobile telephone handset may include a phase locked loop as described. An integrated circuit may include a phase locked loop as described. 
   The invention uses a variety of commonly available elements to derive a frequency synthesiser based system optimised for phase noise and lock time. Particular attention has been given to optimising the application of each element within the system so they serve their purpose without duplication. This invention relates to a variety of systems all using a digital accumulator core. The second part of this invention relates to using digital dividers to offset some of the limitations of this technique. The applicability of these dividers and their obvious advantages applies equally to other existing applications of direct digital synthesisers. 
   This invention offers both static phase coherency and dynamic phase adjustment. 

   
     BRIEF DESCRIPTION OF THE FIGURES 
     The invention will now he described with reference to the accompanying Figures, in which: 
       FIG. 1  illustrates a prior art phase locked loop; 
       FIG. 2  illustrates a prior art direct digital synthesiser; 
       FIG. 3  illustrates a simple phase locked loop in accordance with the present invention; 
       FIG. 4  illustrates a mix-down phase locked loop in accordance with the present invention; 
       FIG. 5  illustrates the phase locked loop of  FIG. 3  with compensation; 
       FIG. 6  illustrates the phase locked loop of  FIG. 4  with compensation; 
       FIG. 7  illustrates the phase locked loop of  FIG. 3  with a digital divider; 
       FIG. 8  illustrates the phase locked loop of  FIG. 4  with a digital divider; 
       FIG. 9  illustrates a combination of the phase locked loops of  FIGS. 5 and 7 ; 
       FIG. 10  illustrates a combination of the phase locked loops of  FIGS. 6 and 8 ; 
       FIG. 11  illustrates a direct digital synthesiser phase locked loop with a digital divider according to the present invention; 
       FIG. 12  illustrates a direct digital synthesiser mix-down phase locked look with a digital divider according to the present invention; 
       FIG. 13  illustrates a phase locked loop with a complex digital phase frequency detector according to the present invention; and 
       FIG. 14  illustrates a mix-down phase locked loop with a complex digital phase frequency detector according to the present invention. 
   

   DESCRIPTION OF PREFERRED EMBODIMENTS 
   Referring to  FIG. 3  there is illustrated a first example implementation of the present invention in a simple phase locked loop. The simple phase locked loop refers to the PLL  10  of  FIG. 1 . In accordance with the present invention, there is additionally provided a digital accumulator  26  and a phase locked reference clock source  28 . The digital accumulator  26  additionally receives frequency input word  30 . A single bit of the digital accumulator  26 , the most significant bit MSB on line  32 , forms the input to the PLL  10 . The input to the PLL  10  is the reference input to a single bit, digital phase frequency detector  2 . The required output frequency is generated on an output line  36  of the PLL  10 . It should be noted that the digital accumulator  26  and reference clock  28  of  FIG. 3  correspond to the digital accumulator  14  and reference clock  24  of  FIG. 2 . 
   Referring to  FIG. 4 , there is shown a further example of the implementation of the present invention in a mix-down phase locked loop. Referring to  FIG. 41  the mix-down phase locked loop includes the phase frequency detector  2 , low pass filter  4  and VCO  6  of the simple phase locked loop of  FIG. 1 . The divider  8  of the phase locked loop of  FIG. 1  is replaced by an IF filter  38  and a mixer  40 . The feedback input to the phase frequency detector  2  is provided from the required output frequency  36  via the mixer  40  and the filter  38  in series. A further input to the mixer  40  is provided on line  52  from a further voltage controlled oscillator  42 . A reference signal  54  provides an input to a divider  48  having a divider ratio of 1/R. The output of the divider  48  forms an input to a phase frequency detector  46 . The output of the phase frequency detector  46  is presented to a low-pass filter  44 , which drives the VCO  42  to generate the signal  52 . The signal  52  is additionally fed through a divider  50  having a divider ratio of 1/N to provide the second input to the phase frequency detector  46 . The signal on line  52  additionally provides a clock signal to the digital accumulator  26 , which is configured in the same way as the digital accumulator  26  of  FIG. 3 . Thus in  FIG. 4  the digital accumulator  26  receives a digital frequency input word  30  and generates a single signal providing the most significant bit stored in the digital accumulator  26 , on line  32 . 
   The synthesisers described herein in accordance with the invention with reference to  FIGS. 3 and 4  use a digital accumulator to provide a correct frequency-interpolated output signal through the most significant bit of the digital accumulator  26 . This “most significant bit” signal MSB contains the correct mean frequency as well as a strong unwanted phase modulation, since the remainder of the phase information remains untouched in the phase accumulator. Normally, a direct digital synthesiser uses the “P” most significant bits available in the digital phase accumulator to drive the remainder of the device, preserving much of the available phase information. 
   With this simple scheme as illustrated in  FIGS. 3 and 4  there is a problem associated with reducing the strong spur energy found at different, well defined input clock to output frequency ratios. This problem is due to the single most significant bit value only containing the phase values of 0° or 180° necessary to drive the single bit, digital phase frequency detector. To overcome this problem, in a preferred embodiment as described herein below, with reference to  FIGS. 5 and 6 , the invention uses a combination of digital dividers and digital-to-analogue converters to compensate for this excess spur energy. 
     FIGS. 5 and 6  illustrate how the digital phase information available in the digital accumulator may be directly fed into a digital-to-analogue converter. The phase information may also be sub-sampled and fed into a digital-to- analogue converter to reduce the digital-to-analogue converters operating frequency at higher accumulator operating frequencies. 
   Referring to  FIG. 5 , there is illustrated an example implementation of the simple phase locked loop of  FIG. 3  according to the invention with a further modification to include phase compensation. Thus the arrangement of  FIG. 5  further includes a latch  62 , a digital-to-analogue converter  60  and, as will be further described herein after, a divider  68 . A digital word is output from the digital accumulator on line  64 , representing the phase of the digital word stored in the digital accumulator  26 . The output digital word is latched into a latch  62 , which on its output provides a word  66  which forms an input to the digital-to-analogue converter  60 . The digital-to-analogue converter  60  provides an analogue signal on line  74  representing the phase of the digital word stored in the digital accumulator  26 . This analogue signal is provided to a summing unit  72  which removes the phase value from the signal at the output of the phase frequency detector  2 . 
   The latch  62  and digital-to-analogue converter  60  may be clocked directly by the clock signal on line  34  generated by the reference clock  28 . However in a preferable implementation of the invention, and as discussed in further detail herein below, the clock signal on line  34  may be divided by the divider  68  to generate a reduced clock on line  70  for clocking both the latch  62  and the digital-to-analogue converter  60 . 
     FIG. 6  illustrates an example implementation of the mixed-down phase locked loop of  FIG. 4  according to the present invention with phase compensation included. The modification to  FIG. 6  relative to  FIG. 4  is identical to the modification of  FIG. 5  relative to  FIG. 3 , and therefore a detailed explanation to the modifications is not given here. It will be apparent from referring to  FIG. 6  that the latch  62 , digital-to-analogue converter  60 , summing unit  72 , and divider  68  are introduced into the circuit of  FIG. 4  in exactly the same manner as they are introduced into the circuit of  FIG. 3  to provide phase compensation. 
   There are some integer frequencies at which the digital information being passed into the digital-to-analogue converter are not able to correctly compensate for the modulation available on the “MSB” data bit. To overcome this problem the divider  68  feeding the digital-to-analogue converter is adjusted for those frequencies. 
   The preferable lower frequency digital-to-analogue converter  60 , clocked by reduced clock signal  70 , serves to sub-sample the Phase information available in the digital accumulator and apply this information as a correction to the output of the phase frequency detector  2  in the analogue phase locked loop. For lower frequency applications, the digital-to-analogue converter need not sub-sample the accumulator&#39;s contents and the accumulator contents may be fed directly into the digital to analogue converter and the latch  62  and the divider  68  not utilised. 
   The data output from the sub-sampled digital accumulator  26  may, in a further modification, be fed through a simple passive-shaping look up table or active noise shaping element, to reduce the spurious energy parasitic to the digital to analogue conversion operation. This noise shaping entity could also use the most significant bit MSB as one of it&#39;s inputs. 
   One distinct advantage of digital accumulators is their inherent ability of produce absolute frequencies with frequency resolutions limited only by the accumulator length. Therefore as a means of minimising spur problems adjustable digital dividers can be additionally used to divide down the master clock frequency into the accumulator, as is illustrated by  FIGS. 7 and 8 . 
   Referring to  FIG. 7  there is illustrated the simple phase locked loop of  FIG. 3  incorporating such an additional divider. As can be seen from  FIG. 7 , a digital divider  80  is introduced between the reference clock  28  and the clock input of the digital accumulator  26 . Thus a divided clock signal is provided by the digital divider  80  on line  84  from the reference clock signal on line  34 . Referring to  FIG. 8  there is similarly illustrated a modification to the mixed-down phased locked loop of  FIG. 4  to include an additional divider. Again, the digital divider  80  is provided to divide down the clock signal  52  to provide a reduced clock signal  82  to the digital accumulator  26 . 
   It is necessary for the frequency word  30  programmed into the digital accumulator  26  to be modified to accommodate a division value D in the digital divider  80  of  FIGS. 7 and 8  and still provide a final output frequency. 
   The digital divider  80  effectively changes the clock frequency driving the accumulator  60 , thereby altering the positions of the inter-multiplication products responsible for spurious products. This technique enables the strongest of these products to be avoided, but does not remove them in all cases. The modification to the formula given earlier above is: 
   
     
       
         
           
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   Where “D” is the preceding reference clock division value, i.e. the division value of the divider  68 . 
   Using the digital divider  80 , has the advantage of reducing the overall clock frequency of the digital accumulator  26  whilst still reducing the stronger spur levels of higher frequency accumulators. This is made possible because the divider  80  can be adjusted for each frequency to ensure the stronger accumulator spur “hot spots” are avoided. 
   Reducing the clocking frequency of the digital accumulator has the additional distinct advantage of reducing the overall system power during normal operation. 
   The advantageous divider as illustrated with reference to  FIGS. 7 and 8  can be additionally and advantageously included in the phase compensation arrangements for  FIGS. 5 and 6 . Referring to  FIG. 9 , there is illustrated a preferable implementation of the simple phase locked loop arrangement with phase compensation as shown in  FIG. 5  further incorporating the digital divider as shown in  FIG. 7 . As can be seen from  FIG. 9 , the effect of the digital divider in such an arrangement is to reduce the clock signal not only to the digital accumulator  26 , but additionally to the latch  62  and to the digital-to-analogue converter  60 . 
   Referring to  FIG. 10 , there is similarly shown the introduction of the digital divider  80  of  FIG. 8  introduced into the mix-down phase locked loop with compensation as shown in  FIG. 6 . As for  FIG. 9 , the digital divider  80  results in a reduced clock signal being applied to the digital accumulator  26  as well as the latch  62  and digital-to-analogue converter  60 . It will be apparent from the foregoing description and from referring to  FIGS. 9 and 10 , that the clock signal applied to the latch  62  and the digital-to-analogue converter  60  is further reduced by the divider  68  when the divider  68  is utilised. 
   The position of the frequency spurs bears a simple relationship with the master clock (ie the reference clock  28 ) frequency, which can be calculated using the accepted inter-modulation formula:
 
Spur Frequencies=± N×F   Accumulator   ±M×F   Digital Equivalent Output Frequency 
 
   Where N and M are integers. 
   What are the integers N and M—how are they derived/determined. The integers n and m range from negative to positive offsets. For example, n is kept at a particular integer value within the offsets, whilst m is varied across the offsets. This is done for every value of n, and in this way a graphical table of values can be established. In this way, the above equation may be used to predict where spur energies will fall for any given output frequency. 
   The principle of utilising a digital divider to reduce the clock frequency of the digital accumulator  26  and thereby minimise frequency spurs can also be applied to direct digital synthesisers. Referring to  FIG. 11 , there is illustrated the simple phase locked loop with digital divider as shown previously in  FIG. 7  further adapted to include a direct digital synthesiser for generating the input to the phase frequency detector  2 . Referring to  FIG. 11 , the digital accumulator  26  generates an output word  104  which forms an input to sine look-up table  100 . The sine look-up table generates an output word  106  to a digital-to-analogue converter  102  which provides an analogue signal on line  108  to the phase frequency detector  2 . The implementation of such a direct digital synthesiser utilised in the digital accumulator  26 , the sine look-up table  100 , and the digital-to-analogue converter  102  is well known in the art. 
   Similarly referring to  FIG. 12 , the mix-down phase locked loop including the additional divider for reducing the clock signal to the digital accumulator  26  of  FIG. 8  is adapted as shown in  FIG. 12  to include the sine look-up table  100  and the digital-to-analogue converter  102  to generate the signal on line  108  to the phase frequency detector  102 . 
   When the output frequency of the digital accumulator  26  is known to fall close to one of the problem frequencies, determined using the expression for spur frequencies stated above, the digital divider  80  is reprogrammed. This requires the digital accumulator  26  to also be reprogrammed to synthesise another frequency which is removed from the vicinity of the known problem frequency. Consequently, the output frequency of the digital accumulator  26  will be correct to drive the phase locked loop, but will possess lower unwanted spur energy. 
   In an alternative arrangement the digital-to-analogue converter  60  of  FIGS. 5 and 6  (or  FIGS. 9 and 10 ) could be replaced (or supplemented) using a suitable digital phase frequency detector which accepts a constant update of (sub-sampled) phase information from the digital accumulator  26 , as part of its operation. 
   Such an alternative arrangement is illustrated in  FIGS. 13 and 14 . Referring to  FIG. 13 , the direct digital synthesiser phase locked loop including the digital divider as shown in  FIG. 11  is modified to include a divider  110  and a complex digital phase frequency detector  112  and the latch  116  in place of the sine look-up table  100  and the digital-to-analogue converter  102  and the phase frequency detector  2 . Referring to  FIG. 13 , the digital accumulator  26  outputs a digital word  104  to a latch  116 , which in turn provides digital word  118  to the complex digital phase frequency detector  112 . In addition the digital accumulator  26  outputs the most significant bit of the word stored therein on line  32  to the divider  110  which provides a divider signal on line  114  to the complex digital phase frequency detector  112 . The digital accumulator  26  is clocked by the reference clock signal on line  34 , the latch  116  is clocked by a divided reference clock signal, provided by the divider  68  dividing the reference clock signal on line  34 . 
   The direct digital synthesiser mix-down phase locked loop of  FIG. 12  may be similarly adapted to include the latch  116 , divider  110 , and complex digital phase frequency detector  112 , as shown in  FIG. 14 . 
   Digital phase information is continuously loaded into the phase frequency detector  112 , where the value is compared to the current phase offset it has determined allowing it to decide an output value. Using the more complex digital phase frequency detector  112 , it is possible to utilise more of the data values available in the digital accumulator without using (or supplementing) the DAC. 
   Important to the invention is the integration of each element&#39;s functionality within the system to avoid unnecessary duplication. Using only the digital accumulator  26  as shown in  FIGS. 13 and 14  removes the high frequency digital-to-analogue converter  102  and sine look-up table  100  (and the optional reconstruction filter), all of which are difficult to implement at high clock speeds. Instead the analogue phase locked loop of  FIGS. 13 and 14  acts like a parametric sine look-up table, high frequency digital-to-analogue converter and tracking reconstruction filter, by translating the phase information available in the digital accumulator  26  through the VCO  6 , within the analogue phase locked loop, to the required output frequency. The analogue phase locked loop acts as a self-oscillating sinusoidal output, as well as a high “Q” factor tracking reconstruction filter. One particular advantage is the output signal of the analogue phase locked loop is not limited to Nyquist sampling theory, as a direct digital synthesiser driven digital-to-analogue converter would be. Thereby an excellent signal-to-noise ratio is preserved. The total system becomes a digital and analogue hybrid circuit, using the easiest equivalent analogue or digital block necessary to complete the system. 
   In each of the above example implementations of the present invention a mix-down approach has been included with the digital divider approach. Using the mix down approach in the analogue phase locked loop removes sufficient closed loop gain reducing the amplification of digital phase/frequency detector noise and any residual spur energy. Therefore, in contrast to the low loop bandwidths used in standard phase locked loops, this loop requires a large loop bandwidth (which is ideal for very fast lock times) to minimise the overall phase noise profile seen at the output. 
   In the examples given hereinabove illustrating the invention being utilised in a mix-down approach, a single PLL is shown providing the clock signal to the accumulator  26  and the mixer  40 . However the invention is not limited to such an arrangement. In one alternative, the respective clock signals provided to the accumulator  14  and the mixer  40  may be provided by separate PLLs, each driven by a common reference signal. 
   For all the above examples the input to the digital accumulator  26  can be used for absolute phase adjustment of the analogue output signal. Alternatively it can be used for phase modulating this output signal to accurately reproduce any phase modulated output at much higher frequencies. This phase modulation capability could also be used to introduce noise shaping into the system to reduce the spur energy still further. Any sigma delta or noise dithering scheme could be employed. 
   It should be noted that the invention has been described herein with reference to particular examples. The invention is limited in its scope by the appended claims, and the applicability of the present invention may be broader than that as discussed above in the examples given.