Abstract:
The invention relates to a filter device ( 341, 342 ) for detecting and/or removing erroneous components like noise, deformations, glitch components or other errors in and/or from a signal, to a demodulation device using the filter device, to an information transmission system using the demodulation device and to a method for detecting a noise impulse in an input signal (Cα(t), Cα[k]). The filter device includes a summing element ( 510 - 51 N,  610 ) connected to a correction element ( 540, 640 ). The summing element ( 510 - 51 N,  610 ) sums the input signal (Cα(t), Cα[k]) within a reference interval (N) and the correction element ( 540, 640 ) verifies the summed input signal (IS[k]) with at least one signal condition ( 0 , N+1, ΔCα[k]). Finally, the correction element ( 540, 640 ) outputs a predetermined signal (Cα α [k]) based on the result of the verification between the summed input signal (IS[k]) with at least one signal condition ( 0 , N+1, ΔCα[k]). The foregoing filter device is able to remove noise from an input signal. This stabilizes the input signal against environmental influences occurring e.g. during signal transmission.

Description:
FIELD OF THE INVENTION 
     The invention relates to a filter device for detecting and/or removing erroneous components in and/or from a signal, to a demodulation device using the filter device, to an information transmission system using the demodulation device and to a method for detecting a noise impulse in a signal. 
     BACKGROUND OF THE INVENTION 
     Complex transmission schemes as used in commonly known transmission systems like mobile communication, radio broadcasting or satellite transmission are well known in the fields of information transmission. Therein, information is complexly transmitted from a transmitter to a receiver locally displaced to each other. In the transmitter, a complex modulator multiplies an information signal with two linearly independent signals. The result is a complex modulation signal including two real signal components. Next, the complex modulation signal is modulated onto a high-frequency signal and transmitted from the transmitter to the receiver via an antenna. In the receiver, a complex demodulator reconstructs the information signal from the complex modulation signal based on a suitable demodulation technique. The complex and high-frequency modulation can also be performed in one single step. Complex transmission schemes were introduced in form of the single side band modulation, which was especially used for information transmission in the high-frequency spectrum (HF-spectrum). 
     The linearly independent signals used for generating the complex modulation signal are usually chosen as sinus-signal and cosinus-signal. Since these signals comprise the same amplitude but a phase-difference of 90°, the complex modulator and demodulator can be described with complex numbers facilitating the overall calculation and design process. The sinus-signal is known as in-phase signal, wherein the cosinus-signal is known as quadrature signal. The main advantage of complex transmission schemes is an efficient utilization of the available bandwidth. 
     Subsidiary, but important transmission schemes within complex transmission schemes are digital transmission schemes. Therein, the information signal includes a bit stream, which can be particularly efficiently transmitted in the above described complex modulation signal. Practically, the bit stream and the complex modulated signal are divided into a plurality of symbols each comprising the same length. Each bit sequence of the same structure will be allocated to a predetermined symbol type having a unique amplitude and phase in the complex modulation signal. Such digital complex modulation schemes are well known in the prior art and provide the advantage, that the information included in the information signal can be transmitted with higher quality compared to analog complex modulation schemes. 
     Particularly of interest in the fields of digital complex modulation schemes are complex angle modulation schemes like phase shift keying (PSK) and frequency shift keying (FSK). Therein, the information signal including the bit stream is merely modulated into the phase of the complex modulation signal. Although these transmission schemes inefficiently utilize the available bandwidth, the main advantage is that the signal is more robust against non-linearities. 
     Complex demodulators for demodulating the above described complex modulation signal may work according to a plurality of technical principles. The most important complex modulator types are cross-differential-cross multiply demodulators (CDM), limiter-discriminator integrate demodulators (LDI) and zero crossing demodulators (ZCD). Whereas a CDM demodulator is highly complicated since it requires a lot of electronic components leading to high power consumption, a LDI demodulator is highly error prone. A good compromise could be provided by the ZCD demodulator combining the advantages of the LDI demodulator and the CDM demodulator. However, this is only given, if the amount of different symbol types in the complex modulation signal is sufficiently high. 
     The ZCD demodulator basically consists of a plurality of phase detectors and a reconstructing processor. Each of the phase detectors receives the complex modulation signal and provides its output to the reconstructing processor. There is one phase detector for each predetermined symbol type detecting whether the phase of the modulated complex signal is higher or lower than the phase of the predetermined symbol type. Based on this information, the processor determines whether the modulated complex signal turns clockwise or anti-clockwise in the complex area and reconstructs the whole bit stream of the information signal based on predetermined mathematics. 
     Theoretically the output of each phase detector is constant at least over the duration of the predetermined symbol type. However, practically in the receiver, the complex modulation signal is noisy. This makes the complex modulation signal jittering while turning. However, due to this jittering, the complex modulation signal crosses the phase of a predetermined symbol type forward and backward with very high frequency. This leads to to a plurality of very short pulses output by the respective phase detectors. These short pulses are glitch components and therewith noise in the received information signal. In other words, the jittering could affect and even distort the whole reconstruction processing after the phase deciders. 
     OBJECT AND SUMMARY OF THE INVENTION 
     It is therefore an object of the invention to provide a filter device for removing deformations, glitch components or other errors from a signal and a method for detecting deformations, glitch components or other errors in a signal. 
     The invention is based on the thought that noise like deformations, glitch components or other errors in a signal cannot be detected at a predetermined point in time but only by analyzing the signal over a complete time period in one piece. Thus, to detect errors in a signal, the invention proposes to define a reference interval, which length is preferably longer than each erroneous component in the signal to be detected. To regard the signal in one piece, it is required to store the signal within the reference interval. This can be best achieved by summing the signal over the reference interval. To detect or remove the erroneous component, the summed signal needs to be verified with a predetermined condition—e.g. the sum of the ideal signal. The result of this verification can thus be taken to either unambiguously determine the presence of an erroneous signal component and/or to immediately correct the input signal. 
     Therefore, the present invention proposes a filter device for removing noise like deformations, glitch components or other errors from an input signal, which may be a received and modulated signal in a transmission system. The filter device includes a summing element and a correction element. The summing element sums the input signal within a reference interval. Based on a verification between the summed input signal and at least one signal condition, the correction element outputs a predetermined signal as filter output signal. In the context of the invention, “predetermined signal” means a signal determined within the correction element by reading out a memory, by calculation or by other suitable determination procedures prior outputting. 
     Due to the summing element, it is possible to catch a measurement value for observing the input signal over the reference interval in one piece. The advantage is that a sum can technically be realized in a simple way and verified with predetermined conditions, which are suitable to detect deformations, glitch components and other errors in the input signal. In other words, due to the observation of the input signal over a reference interval in one piece, an electronic signal can be easily corrected without bulky hardware resources. 
     Firstly, different embodiments for the summing element should be provided. 
     In one embodiment, the summing element may include a buffer element serially connected between an input of the filter device and a summing unit, wherein the buffer element timely delays the input signal. Therein, the input and the output of this buffer element are summed by the summing unit. 
     Alternatively, at least two buffers are connected in series between the input of the filter device and the summing unit. Therein, the first buffer may timely delay the input signal and the at least second buffer element may timely delay the output of its preceding buffer element. The total time delay of all buffer elements together may be less than or equal the reference interval. The summing unit sums the input of the first buffer element and the output of all buffer elements as summed input signal. 
     Using buffers and a summing element for summing the input signal is a suitable embodiment, which can be fully realized by hardware or by software. Since software solutions do not require new electronic components, the filter device can be easily integrated into existing electronic components, which are able to process software. 
     The summing element may be an integrator adapted to integrate the input signal during the reference interval. In the context of this embodiment, an integrator can be realized as analogue component (e.g. a capacitor) or as digital component (e.g. a counter). 
     Since integrators like capacitors and counters are cheap, space-saving and low in power consumption they do not only reliably fulfill their technical task but are also cost-effective and simply to produce. 
     Next, different embodiments for the at least one signal condition should be outlined. It can be roughly separated into likelihood-conditions and identity conditions. Based on a likelihood condition, a measured signal is allocated to one of at least two different ideal signal shapes. In contrary thereto, identity conditions are suitable to determine whether a signal deviates from at least one ideal signal shape. 
     The at least one signal condition may be a threshold level and the correction element may output a first predetermined signal if the summed input signal raises the threshold level, and a second predetermined signal if the summed input signal falls below the threshold level. 
     The threshold level is a likelihood condition. Therein, the noise in the input signal will not be classified by quality, intensity or duration but only removed. This embodiment gets especially effective if the input signal is simple. In case of a binary input signal (the signal has only two levels) the filter process is reduced in so far that only one decision for determining the output signal of the filter device is necessary. Thus, the less decisions are necessary, the less calculation time is required, reducing the response time of the whole filter device. 
     The at least one signal condition may be a first and/or second signal shape condition indicating the summed value of a first predetermined signal and a second predetermined signal within the reference interval. Therein, the correction element may either output the input signal, if the output of the summing element corresponds to the first or second signal shape condition or output a corrected input signal in all other cases, wherein the corrected input signal may be created based on at least one preceding input signal value. 
     The predetermined signal shapes as signal conditions are identity conditions that enable to determine whether the flow of the input signal deviates from an ideal signal flow. In other words, it is not only possible to correct the input signal but also to classify the noise in the input signal in respect to suitable conditions like quality, intensity or duration. This information can be used further to evaluate variables about the source or the transmission channel of the input signal and therewith to facilitate the signal correction. 
     The input signal may be a pulse shaped signal carrying digital information. 
     Pulse shaped signals for indicating digital information are not only easy to sum, the at least one signal condition for the input signal can be formulated also very simple. It therefore allows implementing a less complex filter device with a quick response time and low energy consumption. 
     The pulse shapes of the input signal may be rectangular. If the correction element detects noise, a preceding value of the input signal may be outputted one more time. 
     Since rectangular pulses have a constant pulse level over a predetermined time period and the correction element merely outputs preceding input signal values for correcting the input signal, it is possible to omit a difficult and complex prediction. This further reduces the complexity of the filter device. 
     The reference interval may be longer than a longest noise component in the input signal. 
     By doing this, it is secured, that all noise components can be detected within one reference interval. 
     The present invention further proposes a demodulation device including a filter device as described above. 
     The demodulation device reconstructs an information signal based on a modulated signal including at least one modulated signal component. It includes a separation element, a pre-processor and a processor. The separation element separates the modulated signal into the single modulated signal components (e.g different symbols types included in the modulated signal). The pre-processor includes at least one filter device according to the present invention for each of the modulated signal component and reduces or removes noise from each of the modulated signal components. The filtered modulated signal components are finally reconstructed to an information signal in the processor. 
     Due to filtering the modulated signal components, parasitic pulses are removed, such that the overall robustness of the demodulation device is increased. This allows constructing the demodulation device more robust and fault-tolerant with simple hardware means. This saves production and material costs for the demodulation device. 
     The demodulation device may be a zero crossing detector. 
     Since the phase crossings in a zero crossing detector are very prone to parasitic pulses occurring due to jittering of the modulated signal around the phase of a predetermined symbol type, the present invention is especially suitable for these kinds of demodulation devices. 
     The demodulation device may further including a post-processor for deriving predetermined boundary information having a periodic structure in the information signal. 
     The periodic structure in the modulated signal can be used to define the reference interval or the time limits during which the pre-processor should perform the noise reduction. The boundary information can be easily derived from the information signal itself, based on considering e.g. a bit stream. On contrary thereto, it is technically complex to derive the boundary information from the modulated signal directly. Thus, the boundary information derivation in the pre-processor is not only simple but also technically and economically effective. 
     The boundary information may be the start and end of a bit-sequence in the information signal. 
     This embodiment is suitable, since in digital information transmission, bit-sequences are modulated onto a pulse in the modulation signal. Thus, these bit-sequences are a suitable indicator for the pulses of the modulation signal and therewith for the reference interval. 
     The boundary information may be derived based on a phased-locked loop. 
     If bit transitions occur at equally spaced time intervals, they can be detected in the demodulated signal. If the provided signal contains enough bit transitions (which can be guaranteed with suited data encoding techniques), the bit or symbol boundaries can be recovered using a phase lock loop. 
     A transmission system according to the present invention for transmitting an information signal includes at the transmitting side a modulator, an optional up-mixer and a first antenna for transmitting a transmission signal, and at the receiving side a second antenna for receiving the transmission signal, an optional low-noise amplifier, an optional down-mixer and a demodulator as described above. The modulator at the transmitting side modulates the information signal to a modulated signal, which is optionally mixed with a high-frequency signal to a transmission signal in the up-mixer. In case of a direct carrier modulation system, the up-mixer can be omitted. This transmission signal is transmitted between the two antennas. At the receiving side, the optional low noise amplifier amplifies the received transmission signal and provides it to the optional down-mixer, which optionally removes the high-frequency signal from the received transmission signal in case of a non-direct carrier modulation system. Finally, the demodulator reconstructs the information signal out of the received transmission signal provided by the down-mixer. 
     In a preferred embodiment, the transmission system is a magnetic induction transmission system as used in wireless data transmission, in mobile communication or in Bluetooth communication, to which the present invention can be employed very effectively, since magnetic induction systems are very non-linear and noise-dependent. 
     A method for detecting noise like deformations, glitch components or other errors in an input signal includes summing the input signal within a reference time interval, verifying the summed input signal with at least one signal condition and generating a noise detection signal indicating whether the summed input signal fulfils the at least one signal condition. Therein, the noise pulse has a noise duration shorter than the reference time interval. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       The invention will be described in greater detail hereinafter, by way of non-limiting examples, with reference to the embodiments shown in the drawings. 
         FIG. 1  is a signal transmission system including a filter device according to the present invention; 
         FIG. 2  is diagram discussing the H-field from the transmitter antenna of the signal transmission system versus the distance; 
         FIG. 3  is a demodulator included in the signal transmission system according to the present invention; 
         FIG. 4  is a diagram illustrating the working principle of the phase decider included in a zero-crossing demodulator according to the present invention; 
         FIG. 5  is a first embodiment of the filter device according to the present invention; 
         FIG. 6  is a second embodiment of the filter device according to the present invention. 
     
    
    
     DETAILED DESCRIPTION OF EMBODIMENTS 
       FIG. 1  is a signal transmission system including a filter device according to the present invention. The signal transmission system  100  includes a transmitter  110 , a transmission path  120  and a receiver  130 . Therein, the transmitter  110  operates according to a conventional transmitter and modulates an information signal onto a transmission signal, which will be transmitted via the transmission path  120  to the receiver  130 . The receiver  130  finally reconstructs the modulation signal from the received transmission signal and provides the information signal based on the reconstructed modulation signal. As described later in further detail, the transmission path  120  distorts the transmission signal by noise, such that the receiver  130  does not receive an exact copy of the original transmission signal. However, this distortion influences the reconstruction of the information signal. With the filter device according to the present invention, the noise in the received transmission signal or the reconstructed modulation signal can be reduced or even completely removed. 
     In a preferred embodiment, the transmitter  110  includes a modulator  111 , an up-mixer  112 , a coil driver  114  and a first antenna  115 . The modulator  111  takes the information signal and creates based thereon the modulation signal. This modulation signal is then mixed with a high-frequency signal in the up-mixer  112  to create the transmission signal, which is finally provided to the coil driver  114 . The coil driver  114  prepares the transmission signal by e.g. amplifying and transmits it via the first antenna  115  to the receiver  130 . 
     The receiver  130  preferably includes a second antenna  135 , a front end  134 , a low-noise amplifier  133 , a down-mixer  132  and a demodulator  131 . The frond end  134  receives the transmission signal on the frequency of the high-frequency signal via the second antenna  135 . The low-noise amplifier  133  amplifies the received transmission signal and provides it to the down-mixer  132 , which removes the high frequency signal from the received transmission signal. Thus, in the down-mixer  132 , the modulation signal is reconstructed. Finally, the demodulator  131  reconstructs the information signal based on the reconstructed modulation signal. For facilitating the reconstruction of the information signal, the demodulator  131  includes the filter device according to the present invention. It enables to stabilize the reconstructed modulation signal and to improve the reconstruction of the information signal. 
     Prior discussing the working principle of the present invention in further detail, a model for the transmission path  120  should be discussed in further detail. 
       FIG. 2  is diagram discussing the H-field around the first antenna  115  of the signal transmission system  100  versus the distance. In the diagram, it is assumed that the signal transmission system  100  is based on magnetic induction technology. Thereafter, the first antenna  115  is a coil through which a sinusoidal current representing the transmission signal is flowing to generate a magnetic flux in a quasi-static magnetic field. The second antenna  135  is also a coil, through which the magnetic flux generated from the first antenna  115  is passing and inducing a modulated current representing the received transmission signal. 
     The quasi-static magnetic field generated by the first antenna  115  can be divided into three basic components, a linear inverse term r −1 , a square inverse term r −2  and a cube inverse term r −3 . 
     Square and cube inverse terms r −2 , r −3  are called “near field term” and used to calculate matching pairs of E and B vectors standing orthogonal to each other and to the radial vector of the first antenna  115 . The linear inverse term r −1  is called “far field term” used to calculate both, the E and B vectors, at distances much greater that the wavelength. 
     At short distance from the current loop, near field term dominates and is the major contributor. The cube inverse term r −3  is used to calculate the magnetic field component. It is independent of frequency, such that any frequency can be employed in the near-field domain, for the current in the first antenna  115 , to generate a specified magnetic flux through the second antenna  135  in the receiver  130 . In the near-field region of the first antenna  115 , the properties of the quasi-static magnetic field are primarily determined by the source characteristics, and the electric field component is much weaker than the magnetic field component. The total power radiated by the first antenna  115  is however frequency dependent and proportional to λ 2 , wherein λ is the wavelength of the transmission signal transmitted from the first antenna  115 . 
     At the distance of λ/2π from the first antenna  115 , the linear, square and cube inverse terms r −1 , r −2 , r −3  equally contribute to the quasi-static magnetic field. This distance is often referred to as “near field—far field boundary”. 
     At distances larger than λ/2π from the first antenna  115 , the far-field components dominate, the electric and magnetic field components are directly proportional to one another, and the properties of the quasi-static magnetic field depend primarily on the characteristics of the medium through which the quasi-static magnetic field is propagating. 
       FIG. 3  is a demodulator included in the signal transmission system. The demodulator  131  preferably consists of a mixing element  310 , a baseband element  320 , a deciding element  330 , a pre-processing element  340 , a processing element  350  and a post-processing element  360 . The mixing element  310 , the baseband element  320  and the deciding element  330  can be summarized as separation element for separating at least one signal component from a modulation signal. 
     The mixing element  310  receives a modulation signal x(t) and generates a in-phase signal component I(t) and a quadrature signal component Q(t) based thereon. These components are provided to the baseband element  320  generating a baseband signal mα(t) turning in the complex area. Based on the baseband signal mα(t), the deciding element  330  generates crossing signals C∝(t) indicating whether the phase of the baseband signal mα(t) is larger or smaller than a predetermined phase. In the present embodiment these predetermined phases are 0° and 90°. The crossing signals C∝(t) are next provided to the pre-processor  340  for sampling. Further, the pre-processor  340  removes noise from the crossing signals C∝(t) according to the principle of the present invention. The sampled and filtered crossing signals C∝α[k] are provided to the processor  350  generating a pulse sequence p[k] indicating the number of phase crossings of the baseband signal mα(t) in positive and negative direction in the complex area. In view of the present embodiment, this means that the pulse sequence p[k] indicates whether and how often the baseband signal mα (t) crossed the phases 0° and 90° clockwise or anti-clockwise. Finally, the pulse sequence p[k] is provided to the post-processor  360  reconstructing the information signal in form of a bit stream b[n] out of the pulse sequence p[k]. Further, the post-processor  360  derives boundary information l[k] for facilitating the pre-processing in the pre-processor  340 . 
     The mixing element  310  includes two orthogonal signal sources  311 ,  312 , with which the modulation signal x(t) is respectively multiplied. This results into a modulated in-phase signal m I (t) and modulated quadrature signal m Q (t) each comprising a high frequency and a low frequency portion. The low frequency portion is respectively filtered by low-pass filters  313 ,  314  resulting into the in-phase signal component I(t) and the quadrature signal component Q(t). 
     The baseband element  320  includes a mixing element  321  mixing the in-phase signal component I(t) and the quadrature signal component Q(t) together to the baseband signal mα(t). 
     In the present embodiment, the deciding element  330  includes a first decider  331  for outputting a first crossing signal C 0 (t) depending on whether the phase of the baseband signal mα(t) is smaller or larger than the 0° and a second decider  332  for outputting a second crossing signal C 90 (t) depending on whether the phase of the baseband signal mα(t) is smaller or larger than the 90°. 
     The pre-processor  340  includes a first and second filter device  341 ,  342  according to the present invention for respectively filtering and sampling the first crossing signal C 0 (t) and second crossing signal C 90 (t). Both filter devices  341 ,  342  may further receive the boundary information l[k] for receiving information about the limits of the bit sequence b[n] to be demodulated by the demodulator  131 . 
     The processor  350  includes two delay elements  351 ,  352 , three summers  353 ,  354 ,  357  and two multipliers  355 ,  356 . These elements together are used to realize the following equation:
 
 p[k]=C   0   [k]·C   90   [k −1] −C   0   [k −1] ·C   90   [k]   (1)
 
     This equation represents the basic principle for a zero-crossing demodulator enabling to detect whether and how often the baseband signal mα(t) crosses one of the two phases 0° and 90° clockwise or anti-clockwise. In case of clockwise crossing, a positive pulse will be generated based on equation (1). In case of anti-clockwise crossing, a negative pulse will be generated based on equation (1). As well known for a skilled person, a detection in a zero-crossing demodulator can be performed based on more than two predetermined phases. Thus, in a zero-crossing demodulator, equation (1) can also be applied to more predetermined phases standing orthogonal to each other in the complex area, such that the detection of the movement of the baseband signal mα(t) can be performed arbitrarily exactly with a suitable sensitivity. A further detailed explanation of the zero crossing demodulation principle will be given later. 
     The post-processor  360  includes a counter  361  and a phased locked loop  362 . The counter  361  counts the pulses in the pulse sequence p[k] and generates the bit stream b[n]. The generation of the bit stream b[n] based on the pulse sequence p[k] is a well known procedure and should not be explained in further detail. As for example, in the case of FSK modulation, the bit stream b[n] can be recovered by counting when the pulse sequence p[k] is sampled. However, if the pulse signal p[k] is analog, the bit stream b[n] can be recovered by integrating a pulse over a bit period. In every case, if the integration or counting result is negative, the output bit of the bit stream b[n] should be a ‘1 ’. Otherwise it should be a ‘0’. Further, the phased locked loop  362  derives the required bit period as boundary information based on the pulses to indicate the limits of each pulse facilitating the sampling and filtering of the crossing signals C 0 (t), C 90 (t) in the pre-processor  340 . 
       FIG. 4  is a diagram illustrating the principle of the pulse generation in a zero crossing demodulator. The diagram shows the complex area, wherein two phases 0° and 90° and their respective linear dependent phases −0° and −90° are indicated. The baseband signal mα(t) is shown by its quadrature components Q(t), I(t) at two different points in time t, t+t 0 . Between these two different points in time, the baseband signal mα(t) crosses the phase 0° clockwise. 
     This phase crossing will be detected as follows. In the deciding element  330  at the time point t, the first decider  331  will output a negative first crossing signal C 0 (t) and the second decider  332  will output a negative second crossing signal C 90 (t), since the phase of the baseband signal mα(t) is smaller than 0° and 90°. Thus, the sampled values for the crossing signals C 0 (t), C 90 (t) at the time point t will be C 0 [k]=−1 and C 90 [k]=−1. After the time t 0 , the baseband signal mα(t) moved, the first decider  331  will now output a positive value, since the phase of the baseband signal mα(t) is now larger than 0°. The output of the second decider  332  will not change. Therefore, the sampled values for the crossing signals C 0 (t+t 0 ), C 90 (t+t 0 ) at the time point t+t 0  will be C 0 [k+1]=1 and C 90 [k+1]=−1. Based on the calculation scheme provided by equation (1), the processor  350  will calculate and output:
 
 p[k +1]=(1)·(−1)−(−1)·(−1)=−2   (2)
 
     Thus, based on the rule provided above (negative pulse=clockwise turning, positive pulse=anti-clockwise turning), the pulse sequence p[k] indicates that the baseband signal mα(t) turns clockwise during the time points t and t+t 0 . 
     However, the deciders  331 ,  332  in the deciding element  330  output analog crossing signals C∝(t). These analog crossing signals C∝(t) are prone to noise especially if the baseband signal mα(t) jitters around the phases 0° or 90° due to noise in the baseband signal mα(t). This would lead to noise pulses in the crossing signals C∝(t). These noise pulses are called parasitic pulses reducing the robustness of the whole zero crossing demodulator. Therefore, the filter device according to the present invention is especially suitable to increase the robustness of a zero crossing demodulator. This filter device can be preferably realized according to the two embodiments hereinafter shown. 
       FIG. 5  is a first embodiment of the filter device according to the present invention. Thereafter, the filter device  341 ,  342  includes a plurality of summing units  510 - 51 N, a plurality of serially connected buffers  521 - 52 N, a sampling element  530  and a correction element  540 . 
     The sampling element  530  samples the incoming crossing signal C∝(t) according to a predetermined sampling frequency f s . The sampled values C∝[k] are stored into the first buffer  521  of the plurality of the buffers  521 - 52 N. Synchronously, the content of the first buffer  521  is moved to the second buffer, and so on. The input of the first buffer  521  and the output of all buffers  521 - 52 N is summed together by the plurality of summing units  510 - 51 N to create an intermediate signal IS[k]. Based on this intermediate signal, the correction element decides whether to output the content of the first buffer  521  or the content of the second buffer as filtered crossing signal C∝α[k]. 
     The decision is made based on the following thought. The incoming crossing signal C∝(t) includes pulses indicating whether the baseband signal mα(t) is higher or lower a predetermined phase. Due to the principle of the used modulation scheme, phase crossings can only occur after predetermined time periods. It is therefore proposed to choose a reference interval, which is lower than these predetermined time periods during which the shape of the incoming crossing signal C∝(t) should be constant. Since the incoming crossing signal C∝(t) is sampled, this reference time interval is chosen as a numeric value N. Now, if the summed sampled incoming crossing signal C∝(t)—namely the intermediate signal IS[k]—is equal to one of the ideal signal shapes, the filter device should output the currently sampled crossing signal value C∝[k]. Otherwise, the filter device should start a correction procedure based on a preceding sampled crossing signal values C∝[k−1], C∝[k−2]. In the simplest case, this correction procedure comprises to output the content of the first preceding sampled signal crossing value C∝[k−1] as indicated in the present embodiment. 
       FIG. 6  is a second embodiment of the filter device according to the present invention. Thereafter, the filter device  341 ,  342  includes an integrator  610 , a sampling element  630  and a correction element  640 . 
     The integrator  610  integrates the incoming crossing signal C∝(t) over a reference interval. According to the considerations taken out in the first embodiment, this reference interval should be shorter than the time periods between the phase crossings of the baseband signal mα(t). The integrated incoming crossing signal C∝(t) of the integrator  610  is sampled by the sampling element  630 , such that the output of the sampling element  630  represents the intermediate signal IS[k] analog to the first embodiment. In case of that the incoming crossing signal C∝(t) is already a discrete signal, the integrator  610  and the sampling element  630  can be substituted by a counter, which directly outputs the intermediate signal IS[k]. In the present embodiment, the correction element  640  decides whether the intermediate signal IS[k] raises a predetermined threshold level ΔC∝. Based on this threshold level, the correction element  640  outputs a first or a second corrected crossing signal value C∝α[k]. 
     Thus, independently of the application of the filter device according to the first or second embodiment, the pre-processor  340  always outputs a noise free crossing signal C∝α[k]. This increases the robustness of the zero-crossing detection and facilitates the overall demodulation procedure. 
     In other words, the present invention provides a filter device applicable in mobile communication, radio broadcasting or satellite transmission and in the fields of information transmission. The filter device allows removing noise from an input signal by summing the input signal, such that the robustness of the input signal is increased by simple means.