Abstract:
The filter for eliminating a carrier signal from a modulating signal generated by a circuit supplying a load (Z) comprises a filtering circuit (L, C; L′, C′) and a cancellation network (Lx, Cx, V 2 ; Lx, Cx, Lx′, Cx′) which, when the load is connected to said filter, subtracts from the load a current equal to the residual carrier signal passing through said filtering circuit (L, C; L′, C′).

Description:
FIELD OF THE INVENTION 
     The present invention relates to an output filter for eliminating the switching residue of a carrier signal from a modulating signal, for example in order to eliminate the switching residue from the signal supplied by a generator, for example a PWM generator, to a load connected to said system. 
     BACKGROUND OF THE INVENTION 
     In particular, in the sector of the audio band amplifiers there are currently used, in addition to linear amplifiers, switching devices which can be schematically described as a pair of switches controlled at a high switching frequency which switch the current to the load in an alternating way. The timing of the switching instants follows a modulation criterion according to the signal to be reproduced. The output signal, consisting of the sum of the carrier signal and the modulating signal, is demodulated in a low-pass filter which extracts the low-frequency modulating signal from the high-frequency carrier. A diagram of an amplifier and of a corresponding low-pass filter of this type which are currently known is provided in FIG.  1 . The numbers  1  and  3  indicate two electronic switches controlled by a PWM circuit indicated in a general way by  5 . The chopped signal at the output is sent to a low-pass filter formed by a single L-C network comprising an inductance L and a capacitance C. The L-C network has a cut-off frequency of the order of 25 kHz to eliminate the high-frequency switching signal from the signal sent to the load Z, so that a signal containing only the frequencies in the audio range is supplied to the load (for example a loudspeaker). 
     With this system, attenuations of 12 dB/octave are obtained. To obtain greater attenuations, it is possible to dispose two or more L-C networks in cascade connection, with a consequent increase of the output impedance of the amplifier. 
     The system may also be of the balanced type, in other words with a bridge configuration of the type illustrated in FIG. 2, where the load Z is disposed between two pairs of controlled switches  1 ,  3  and  1 ′,  3 ′. The control signals of the two pairs of switches  1 ,  3  and  1 ′,  3 ′ are equal but in opposite phase. The symbol L, C and L′, C′ indicate the components of the low-pass output filters. 
     By comparison with amplifiers of the linear type, switching amplifiers have the advantage of a higher theoretical efficiency, equal to 100%, since there are only two possible conditions of the switches, namely: closed switch, zero voltage, non-zero current, zero power dissipation; and open switch, non-zero voltage, zero current, zero power dissipation. 
     At the present time, however, this technology has considerable problems, particularly for application to the audio sector, arising from the limits of the output reconstruction filter which, in order to attenuate the switching residue, also has drastic effects in the pass band, with natural resonances, distortion, and an increase in output impedance at high frequencies beyond the acceptable limits. The limits imposed by the output reconstruction filter are principally as follows: 
     a) increase of the output impedance of the filter as a function of the number of networks and the decrease of the cut-off frequency of each network; 
     b) high variation of the frequency and phase response characteristics as a function of the load impedance; 
     c) insufficient attenuation of the switching residue in cases in which the ratio between the pass band and the switching frequency is high. This problem arises whenever a compromise between quality and switching losses is sought; 
     d) increase of the phase rotation of the output signal in proportion of the number of L-C networks used; 
     e) increase of the resistive part of the filter with an increase in the number of L-C networks which leads to an increase in the conduction losses in the filter and consequently a loss of efficiency; 
     f) increase of the volume and total cost of the filter with an increase in the number of L-C networks for a given switching frequency; 
     g) resonances in the pass band which are dangerous for the operation of the amplifier in no-load conditions. 
     SUMMARY OF THE INVENTION 
     The purpose of the present invention is to provide an output filter for a switching system of the type mentioned above which overcomes the disadvantages of conventional filters. 
     The solution proposed by the invention consists in configuring the output filter on the basis of an L-C network, with attenuation and phase characteristics in the pass band that are acceptable in terms of quality, and adding to this a dipole or cancellation network which provides a very high attenuation at the switching frequency and a minimum effect in the pass band. In practice, the dipole generates a signal equal and opposite to the switching residue at the terminals of the load, in such a way that the sum of the two signals is canceled and no current at the switching frequency flows to the load. 
     The resonant frequency of the L-C network is lower than the switching frequency, so that it presents an inductive behavior to the generator. 
     Further advantageous characteristics of the device according to the invention are indicated in the attached claims and will be described in greater detail with reference to some diagrams showing examples of embodiments. The invention will be more clearly understood from the description and the attached drawing, which shows non-restrictive practical examples of the invention. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIGS. 1 and 2 show two configurations according to the known art as described above; 
     FIG. 3 is a block diagram of the system according to the invention; 
     FIG. 4 shows a solution in a bridge configuration; 
     FIGS. 5 and 6 show successive improvements of the system shown in FIG. 4; and 
     FIG. 7 is a diagram provided for guidance, reproducing a possible response profile of the filter according to the invention. 
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENT 
     With reference to FIG. 1, the concept on which the present invention will initially be illustrated in a particularly simple configuration. The symbol Z indicates in a general way the load to be supplied and L and C again indicate, as in FIG. 1, the inductance and capacitance of the L-C network of the filter. The symbol V 1  indicates schematically the signal source, which contains both the modulated signal and the switching signal. The number  11  indicates in a general way the cancellation dipole connected in parallel to the load Z. It comprises a generator V 2  and an impedance X, consisting of an inductance Lx and a capacitance Cx in series. Additionally, in the diagram in FIG. 3, the symbols I 1 , I 2  and I 3  indicate, respectively, the current flowing in the inductance L, the current flowing in the inductance Lx and the current in the load Z. The reconstruction filter performs its function correctly if: 
     I 2 =−I 1  at the switching frequency w c    
     I 2  approximately 0 in the pass band. 
     If 
     I 2 =V 2 /X=−I 1    
     then the system will have zero voltage across the terminals of the load Z at the switching frequency, with zero current flowing through the load. The correct operation of the filter implies that the value of X is equal to the nominal value V 2 /I 2  at the switching frequency w c  and at the highest possible value (tending towards infinity) in the pass band, in order that the cancellation dipole interferes to the least possible extent with the characteristics determined by the L-C network. In the configuration in FIG. 3, 
     
       
           I   3   =I   1   +I   2 =0; I 1=−I   2    
       
     
     
       
           I   1   =V   1   /jwL  and  I   2   =V   2   j w Cx /(1 −w   2    Lx Cx )  
       
     
     given the condition 
     
       
         V 2 =−V 1 ,  
       
     
     in other words that the two generators are equal and in antiphase, then 
     
       
           V   1   /jwL=V   1    jwCx /(1 −w   2    Lx Cx )  
       
     
     and therefore 
     
       
         1 −w   2    Lx Cx=−w   2    L Cx    
       
     
     from which it is easy to deduce the relationship between the switching frequency w and the values Lx, Cx, and L: 
     
       
           w =1/(( Lx−L ) Cx ) {fraction (1/2 )}   
       
     
     This procedure is formally valid only if the generators V 1  and V 2  are sinusoidal and are mirror images of each other, in other words are in antiphase. 
     As regards the nature of the generators, it is found that the condition of being sinusoidal is not a necessary condition for the final result, since the filter has excellent attenuation, even beyond the switching frequency, of the square wave whose harmonic composition varies as a function of the modulation index. 
     It may be found that multiple alignments of the output filter are possible, since it is possible to keep w c  and L at fixed values and to vary Lx and Cx in such a way as to reach the optimal alignment, in other words the characteristic of attenuation at the harmonics above the switching frequency. FIG. 7 shows an example of the response of the filter: the horizontal axis shows the frequencies and the vertical axis shows the output voltage. The cancellation frequency is indicated by Fc. 
     To meet the condition that the generators should be mirror images of each other, it is preferable to use a balanced configuration, or what is called a bridge configuration, which will be described with reference to FIGS. 4 to  6  and which constitutes the application of the inventive concept illustrated above to a conventional system of the type shown in FIG.  2 . 
     In FIG. 4, the load is again indicated by Z and in parallel to it there is connected a serial RC circuit, consisting of the resistor Rp and the capacitor Cp, to prevent undamped resonances in the absence of the load Z. The load is connected as a bridge between two voltage generators, each consisting of a corresponding pair of driven switches,  1 ,  3  and  1 ′,  3 ′ respectively. The two pairs of switches  1 ,  3  and  1 ′,  3 ′ are driven by the same switching signal but in antiphase. This ensures an optimal symmetry in the two branches which can differ from each other only by the small tolerances between the driving devices and the power switches. These differences, as well as differences between the switching instants of the two pairs of switches, can easily be compensated for as will be indicated in the following text. 
     Between each pair of switches  1 ,  3  and  1 ′,  3 ′ and the corresponding terminals of the load there are connected two L-C networks, one comprising an inductance L and a capacitance C and the other comprising an inductance L′ and a capacitance C′. 
     In order to apply a switching residue cancellation signal to the terminals of the load Z, there is connected, in a similar way to that illustrated in FIG. 3, a first impedance between the first generator consisting of the driven switches  1 ,  3  and the opposite pole of the load Z, and a second impedance between the second generator consisting of the driven switches  1 ′,  3 ′ and the other pole of the load Z. Each of the two impedances consists of an inductance Lx and Lx′ in series with a capacitance Cx and Cx′ respectively. Where the principle of operation and the determination of the parameters Cx, Cx′, Lx and Lx′ are concerned, the disclosures relating to the case shown in FIG. 3 are applicable. 
     The system shown in FIG. 4 is balanced only in a theoretical way, whereas in the practical embodiment the tolerances on the individual components may include differences in the currents flowing in the two branches. In order to equalize the currents in the two branches, the two inductances L and L′ can be coupled by winding them on a common magnetic core. In this case, the value of the overall inductance may be equal to the sum of the two individual inductances L and L′. A number of turnings which is smaller (by a factor of 2 −½ ) than the total number of turnings to be used in the case of non-coupled inductances will therefore be used. This results in a reduction of the ohmic losses in the winding and a reduction in volume, in addition to an advantage in terms of current equalization. 
     The imperfect mirror image configuration of the real system may also be manifested in common-mode disturbances due to the switching residue. The common-mode cancellation is achieved by coupling the inductances Lx and Lx′ by winding them on the same magnetic core. The same considerations are applicable in respect of the number of turnings, and consequently this coupling also permits a reduction of the number of turnings and therefore of the losses in the conductor and of the dimensions. The result is the circuit shown in FIG. 5, where identical or equivalent elements are indicated by the same reference numbers as those used in the diagram in FIG.  4 . The symbol T indicates the winding core of the inductances L and L′, and Tx indicates the winding core of the inductances Lx, Lx′. 
     Additionally, any difference between the switching instances of the two generators  1 ,  3  and  1 ′,  3 ′ can be considered as a phase difference and therefore as a common-mode disturbance in the output signal. This disturbance is eliminated with a common-mode filter which may consist of two coupled inductances as shown in the diagram in FIG. 6, where the common-mode filter consists of two inductances Lm and Lm′, wound on the common core Tm. The remaining components are indicated by the same reference numbers as those used in the diagram in FIG.  5 . 
     Particularly positive results are obtained by using: 
     for the common-mode inductor Lm, Lm′, Tm: high-permeability toroidal cores, in order to have the maximum ratio between inductance and turnings, while minimizing the losses in the conductor (the core does not become saturated since there is no overall flow of a continuous or low-frequency current); 
     for the inductor L, L′, T: low-permeability powdered iron toroidal cores which have very high values of inductance at the saturation, low AC losses at the switching frequency and excellent thermal stability. The use of the magnetic material must remain, in conditions of maximum current to the load, within rather restrictive limits of tolerance in the value of the inductance, to avoid imperfect cancellation of the “ripple”. This condition is present in the materials indicated above; 
     for the inductor Lx, Lx′, Tx: toroidal cores with characteristics similar to those described above, preferably identical in terms of homogeneity of the material and therefore with identical characteristics of the hysteresis cycle. It is preferable to have similar AC induction values (at the switching frequency w c ) in the cores Tx and T in order to optimize the characteristics of symmetry and cancellation. The core Tx is not subjected to high DC values (or those in the pass band) and therefore will have considerably smaller dimensions than the core T of the inductor L, L′, T. 
     It is to be understood that the drawing shows only an example provided solely as a practical demonstration of the invention, and that this invention may be varied in its forms and dispositions without departure from the scope of the guiding principle of the invention. Any presence of reference numbers in the attached claims has the purpose of facilitating the reading of the claims with reference to the description and to the drawing, and does not limit the scope of protection represented by the claims.