Abstract:
A high performance single stage Power Factor Correction (PFC) converter with tight output voltage regulation and a very simple circuit to carry out those functions, which means its cost is lower than its counterparts. Two basic flyback circuits include a simple control circuit. For the hard switching circuit, only one switch is used to achieve low cost; for the soft switching scheme, one auxiliary switch is added to get higher efficiency and smaller size. There are two power flow paths, resulting in part power processed by an active switch only once to reduce the current stress and improve the efficiency. A direct current (DC) bus voltage will be limited to the peak value of input voltage. The maximum DC bus voltage will be less than 400 and a single commercial capacitor can be used for universal voltage stress under light load condition.

Description:
This invention relates to alternating current (ac)/direct current (dc) converter power supplies and more particularly to those which employ a Power Factor Correction (PFC) topology which integrates two flyback topologies enabling a simple control circuit and claims the benefit of priority to U.S. Provisional Application Ser. No. 60/363,145 filed Mar. 8, 2002. 

   BACKGROUND AND PRIOR ART 
   A PFC converter is necessary for many electronic equipments to meet harmonic regulations and standards. For low power applications, single stage PFC converter is a better choice considering cost and performance. 
   Typical single-stage PFC topologies with tight output voltage regulation were proposed in the paper by Redl, R.; Balogh, L.; Sokal, N. O., “A New Family of Single-Stage Isolated Power-Factor Correctors with Fast Regulation of the Output Voltage”, PESC &#39;94, P1137-1144, vol. 2, June 1994. In those topologies, a PFC cell is integrated with a Direct Current/Direct Current (DC/DC) conversion cell, and both cells share active switches and controller, but suffer from high intermediate bus voltage and high current stresses. 
   Some methods to reduce the intermediate bus voltage were discussed in the paper by Qiao, Chongming&#39; Smedley, K. M., “A Topology Survey of Single-Stage Power Factor Corrector with a Boost Type Input-Current-Shaper” APEC 2000, P460-467, Vol. 1, March 2000. Unfortunately, those methods brings high distortion to the line current waveform, resulting in reduced power factor. 
   One approach to limit intermediate bus voltage was proposed in the publication by Luo, et al., “Flyboost Power Factor Correction Cell and Its Applications in Single-Stage AC-DC Converters”, PSEC, 2002. By adding a secondary winding to the boost inductor, there was provided two discharging paths for boost inductor: to the intermediate storage capacitor or directly to the load. It means that some input power is directly transferred to the load without being processed by DC/DC conversion cell, referred as parallel power transfer in both the publications by: Jiang, Y.; Lee, F. C.; Hua, G.; Tang, W., “A Novel Single-Phase Power Factor Correction Scheme”, APEC &#39;93, P287-292, March 1993; and Garcia, O.; Cobos, J. A.; Prieto, R.; Alou, P.; Uceda, J.; “Power Factor Correction: A Survey”, PESC &#39;01, P8-13 vol. 1, June 2001. 
   These approaches limit the intermediate bus voltage with little influence on input current waveform, and allow the DC/DC conversion cell to operate in a continuous conduction mode (CCM) without high voltage punishment at light load conditions. 
   Various patents have been proposed in this area but fail to overcome all the problems of the prior art. 
   A search was also carried out with the following results: 
   U.S. Pat. No. 5,146,394 to Ishii, et al discloses the use of only one power transformer; U.S. Pat. No. 5,796,595 to Cross has two transformers that are not connected in series and is not for Power Factor Correction applications; U.S. Pat. No. 5,909,361 to Kim discloses the use of only one power transformer; U.S. Pat. No. 5,982,638 to Tang, et al discloses the use of only one power transformer; U.S. Pat. No. 6,005,782 to Jain, et al. discloses the use of only one transformer to deliver the energy to the output and is not for Power Factor Correction applications; U.S. Pat. No. 6,031,747 to Ilic, et al, has no transformer, does not work as a Power Factor Correction cell and is not for Power Factor Correction applications; U.S. Pat. No. 6,115,267 to Herbert wherein the transformer does not work in the Flyback mode and is and without an intermediate DC bus capacitor; U.S. Pat. No. 6,272,027 B1 to Fraidlin, et aI has only one transformer and no intermediate DC bus capacitor; U.S. Pat. No. 6,281,666 B1 to Tressler, et al which is not for Power Factor Correction applications and has no intermediate DC bus capacitor; and, U.S. Pat. No. 6,282,109 B1 to Fraidlin, et aI has no isolation, no Flyback transformer nor an intermediate DC bus capacitor. 
   Thus, the need exists for a lower cost, lower bus voltage DC/DC conversion cell that can operate in Continuous Conduction Mode (CCM). 
   SUMMARY OF THE INVENTION 
   It is a primary objective of the present invention to develop a power factor correction topology which limits the intermediate bus voltage to less than 400 volts for universal voltage applications. 
   Another object of this invention is to provide a PFC topology which provides two discharge paths for the choke inductor whereby the intermediate bus voltage has reduced influence on the input current waveform. 
   A further object of this invention is to provide a low cost low power AC/DC converter power supply. 
   A preferred embodiment of the integrates two flyback topologies to enable switching with only a single switch while also requiring only a single capacitor for all universal voltage applications under 400 volts. 
   Further objects and advantages of this invention will be apparent from the following detailed description of presently preferred embodiments which are illustrated schematically in the accompanying drawings. 

   
     BRIEF DESCRIPTION OF THE FIGURES 
     FIG.  1 ( a ) is a schematic of the basic topology of the Bi-flyback PFC converter. 
     FIG.  1 ( b ) schematic of the Bi-flyback PFC converter with a low loss snubber circuit. 
     FIG.  1 ( c ) is a schematic of the bi-flyback PFC converter with an active clamp circuit. 
       FIG. 2  shows the operation mode in one line period. 
     FIG.  3 ( a ) shows the operation waveforms resulting from operation of the flyback mode 
     FIG.  3 ( b ) shows the equivalent circuit during the ON period in operation of the flyback mode 
     FIG.  3 ( c ) shows the equivalent circuit during the OFF period in operation of the flyback mode 
     FIG.  4 ( a ) shows the operation waveforms resulting from operation of the boost mode 
     FIG.  4 ( b ) shows the equivalent circuit during the ON period in operation of the boost mode 
     FIG.  4 ( c ) shows the equivalent circuit during the OFF period in operation of the boost mode 
       FIG. 5  shows the power flow over a line period. 
       FIG. 6  illustrates the input current and input voltage at 100 W output and 110V input. 
       FIG. 7  graphs the measured intermediate bus voltage versus output power. 
       FIG. 8  graphs the measured power factor at a 150 W load. 
       FIG. 9  graphs the measured efficiency at a 150 W load. 
   

   DESCRIPTION OF THE PREFERRED EMBODIMENTS 
   Before explaining the disclosed embodiments of the present invention in detail, it is to be understood that the invention is not limited in its application to the details of the particular arrangements shown since the invention is capable of further embodiments. Also, the terminology used herein is for the purpose of description and not of limitation 
   According to this invention, the above objects can be achieved by a single stage Bi-flyback Power Factor Correction (PFC) topology. The advantages and features of the present invention will be apparent upon consideration of the following description. 
   The Bi-flyback topology consists of two flyback circuits, as shown in FIG.  1 ( a ). The first PFC flyback circuit is composed of transformer T 1   120 , rectifier input bridge  140 , diode D 1    160 , output filter capacitor C o    180  and Power Metal Oxide Silicon Field Effect Transistor (MOSFET) Switch (S)  200 . The second Direct Current/Direct Current (DC/DC) flyback circuit includes transformer T 2   220 , intermediate bus capacitor C S    240 , diode D 2    260 , output filter capacitor C o    180  and the same Power MOSFET Switch (S)  200 . The PFC cell operates under the Discontinuous Conduction Mode (DCM) and the DC/DC conversion cell operates under a Conduction Mode of the class of the Continuous Conduction Mode (CCM) and DCM during the entire line period. 
   The above referenced second DC/DC flyback circuit operates similar to a DC/DC flyback circuit. For the first PFC flyback circuit, there are two discharging paths for transformer T 1   120 , depending on instantaneous input voltage and intermediate bus voltage. When input voltage is low, T 1   120  will discharge its magnetizing energy to the load  280  via diode D 1    160  during S  200  OFF period, like the typical flyback transformer. When input voltage is high, T 1   120  operates like a boost inductor and discharges its magnetizing energy to intermediate bus capacitor  240  through T 2 &#39;s  220  primary winding. 
   When T 1   120  operates like flyback transformer, it is referred to as the flyback mode, and as the boost mode when T 1   120  works as boost inductor. So there are two operational modes for this topology. The operational modes over one line cycle are shown in FIG.  2 . 
   In FIG.  1 ( b ), one low loss snubber circuit, comprised of diode  222  and diode  224 , capacitor  228 , and inductor  226 , is added to basic Bi-flyback topology to absorb the voltage spike across the main switch S when it is turned off. In FIG.  1 ( c ), one active clamp ciruit, comprised of capacitor  232  and MOSFET switch s  234 , is used to carry out snubber function and zero voltage switching for the main switch. 
   A. Flyback Mode 
   When rectified line voltage |V in (t)| is less than V CS +n 2 V o −n 1 V o  (n 1 : the turn ratio of T 1   120 , n 2 : the turn ratio of T 2   220 ), Transformer T 1   120  works like a flyback transformer, and the topology operates like two independent flyback converters. All input power during this mode is directly transferred to the load  280  through T 1   120 . Meanwhile, the DC/DC flyback cell will deliver some power from intermediate capacitor  240  to load  280 , in order to keep tight output voltage regulation. The equivalent circuits and operational waveforms are shown in FIG.  3 . 
   Interval 1 (t 0 ˜t 1 ): Switch S  200  is turned on at t 0 . The rectified line voltage |V in (t)| is applied to primary winding of T 1   120 . The current in T 1   120 , i 1  in  FIG. 3 , increases linearly. The intermediate bus voltage V CS  is applied to primary winding of transformer T 2   220 , causing its current i 2  to linearly increase also. Since PFC cell operates in DCM, i 1  starts increasing from zero.
 
Interval 2 (t 1 ˜t 2 ): S  200  is turned off at t 1 . T 1   120  discharges through its secondary winding and deliveries stored magnetizing energy to the load  280 . The current (i D1 ) in T 1   120  secondary winding decreases linearly. T 2   220  also discharges its magnetizing energy to the load  280  via its secondary winding. The voltage across transformer T 1   120  primary winding is n 1 V o , while the voltage across T 2   220  primary winding is n 2 V o . So the voltage across switch S  200 , V DS , is equal to V CS +n 2 V o , which is higher than |V in (t)|+n 1 V o . The input rectifier bridge  140  is blocked.
 
Interval 3 (t 2 ˜t 3 ): At t 2 , all magnetizing energy in T 1   120  is transferred to the load  280 . Current i D1  reaches zero and diode D 1    160  keeps it at zero. The current in T 2   220  secondary winding, i D2 , continues to decrease until the switch  200  is turned on at t 3 . Then a new switching cycle begins. Switching period T S  is equal to t 3 −t 0 .
 
B. Boost Mode
 
   When the line voltage goes higher, |V in (t)|&gt;V CS +n 2 V o −n 1 V o , the voltage in transformer T 1   120  primary winding during S  200  OFF period is V DS −|V in (t)|, i.e. V CS +n 2 V o −|V in (t)|, which will be less than n 1 V o . It means that diode D 1    160  in T 1   120  secondary winding discharge path will not conduct. T 1   120  works like a boost inductor and discharges its magnetizing energy only through its primary winding. At this mode will operate as BIFRED topology. T 1   120  transfers some input power to intermediate storage capacitor C S    240  and some input power to the load  280  through T 2   220 . Meanwhile, T 2   220  will also transfer some power from intermediate capacitor  180  to the load  280  in order to keep tight output voltage regulation. The equivalent circuits and operational waveforms are shown in FIG.  4 . 
   Interval 1 (t 0 ˜t 1 ): S  200  is turned on at t 0 , resulting in rectified line voltage |V in (t)| applied to T 1   120 . The current in T 1   120  primary winding, i 1  in  FIG. 4 , increases linearly. And the voltage across the intermediate bus capacitor  180 , V CS , is applied to T 2   220  primary winding, which causes the current i 2  in  FIG. 4  to linearly increase also.
 
Interval 2 (t 1 ˜t 2 ): Switch S  200  is turned off at t 1 . T 1   120  current i 1  will decrease linearly and discharge its magnetizing energy through DC/DC transformer T 2   220  primary winding and intermediate capacitor  240 . So T 2   220  primary winding current i 2  is equal to T 1   120  current i 1 , which will be reflected to T 2   220  secondary winding. The current (i D2 ) in T 2   220  secondary winding consists of magnetizing current of T 2   220  and reflected current of T 1   120  current i 1 . And T 2   220  magnetizing current will decrease linearly since output voltage is applied to T 2   220  secondary winding. The voltage across T 2   220  primary winding is n 2 V o . So the voltage across S, V DS , is equal to V CS +n 2 V o . The voltage across transformer T 1   120  primary winding is V CS +n 2 V o −|V in (t)|.
 
Interval 3 (t 2 ˜t 3 ): At t 2 , i 1  reaches zero and the input rectifier bridge  140  prevents it from going negative. And i D2 , which only consists of magnetizing current of T 2   220  in this interval, continues to decrease until the switch is turned on at t 3 . At t=t 3 =t 0 +T S , the switching cycle repeats.
 
C. Features
 
   By adding another discharging path to PFC inductor, the following benefits occur:
         a) The maximum intermediate bus voltage is limited. Only at boost mode when input voltage is higher than V CS +n 2 V o −n 1 V o , intermediate bus capacitor C S    180  is charged by input power. The higher the intermediate bus voltage, the less charging power. So the maximum intermediate bus voltage will be limited to V in,peak +n 1 V o −n 2 V o . Carefully selecting transformer turn ratio n 1  and n 2 , the maximum intermediate bus voltage can be set to a little higher than the peak value of input voltage to achieve low voltage stresses and high power factor. For universal voltage (85˜265V AC,RMS ) applications, the maximum intermediate bus voltage can be controlled to less than 400 VDC, allowing single commercial 450 VDC capacitor to be used in this topology. Since the maximum intermediate bus voltage is limited, DC/DC conversion cell can operate in CCM for low current stresses, without problem of high voltage at light load existing in the conventional single-stage PFC converters.   b) A portion of the load power is processed by the main switch only once. In the flyback mode, all input power is directly transferred to load by T 1   120 . In the boost mode, some input power is directly transferred to the load by T 2   220 , and some input power is stored in intermediate bus capacitor  240  and then delivered to the load  280  by the DC/DC cell. So the total power processed by active switch  200  is less than that in conventional single-stage PFC converter.       

   In  FIG. 5 , the power flows of Bi-flyback topology are illustrated. There are two directly transferred power portions. When instantaneous input voltage is low and Bi-flyback operate under flyback mode, all input power is transferred to output directly by T 1   120 . So P 11  will be equal to input power. When the circuit enters boost mode, some input power is transferred to output by T 1   120 . The sum of the power provided by T 1   120  and T 2   220  should be equal to output power, so the power provided by T 2   220  will be equal to the difference between output power and directly transferred power. 
   Experimental Results 
   One prototype based on the topology shown in FIG.  1 ( c ) has been built and tested to verify its operation principle. The main specifications were:
         Input: approximately 85—approximately 265V AC&#39;RMS   Output: approximately 28 VDC @ approximately 150 W   Switching frequency: approximately 200 kHz   T 1   120 : primary inductance L 1 =30 μH, turn ratio n 1 =4   T 2   220 : primary inductance L 2 =approximately 375 μH, turn ratio n 2 =approximately 3.8   Rectifier Input Bridge  140 : line voltage rectifier rated at approximately 600 V. D 1    160 : approximately 200V/10 Amps fast recovery diode   D 2    260 : approximately 200V/10 Amps fast recovery diode   Power MOSFET s  200 : approximately 600V/10 Amps MOSFET   Intermediate capacitor c s    240 : approximately 450V/approximately 150 μF Aluminum capacitor   Output capacitor c o    180 : approximately 450V/approximately 1000 μF Aluminum capacitor   Load  280 : approximately 150w DC load.       

     FIG. 6  shows experimental waveforms of input current and voltage at 100 W output and 110V AC.RMS input. 
     FIG. 7  shows the variation of intermediate bus voltage versus output power at different input voltage conditions. The maximum voltage across intermediate capacitor is about approximately 390VDC for universal input voltage and al] load condition. 
     FIGS. 8 and 9  gives the measured power factor and efficiency versus input voltage at 150 W load. Measured power factor is approximately 0.974 with approximately 83.2% efficiency at approximately 110V input and approximately 150 W load. 
   It would be useful to now list the key features of the invention which include:
         1. Reduced cost and improved reliability due to least components;   2. Reduced current stress. It can be shown from i L1  in FIG.  3 ( a ), the existence of interval M 3  can effectively reduce the peak current value of main switch when transferring the same average output current. Therefore, the main switch can be turned off under reduced current stress;   3. Reduced voltage stress. Since storage capacitor is charged under the governing equation: V CS &lt;V in (t)+n 1 V o −n 2 V o , therefore, using a flyback transformer to replace the traditional input inductor brings about inherent DC Bus voltage clamping capability. Properly selected n 1  and n 2  can guarantee the DC bus voltage well above the peak value of the input voltage, as a consequence, the commercial available 450VDC capacitor can be used, and moreover, approximately 600V components can be used in power stage for universal input applications;   4. Direct energy transfer. In discharge mode, the input current is directly delivered to the output through the input flyback transformer T 1 . In charge mode, partial power is further directly transferred to the output through T 1  and T 2  without storing in C S  first;   5. Higher efficiency. More than half energy transferred to the load without processed twice undoubtedly can increase the overall efficiency. For the currently existed cascade two-stage or single stage approaches, basically, the power is processed serially by PFC cell and DC/DC cell, the overall efficiency is given by the product of two stage efficiencies, i.e., η=η 1 η 2 , where η 1  and η 2  are the efficiencies of two stages respectively. In the proposed topology, supposing k is the ratio at which power is transferred to the output just through PFC stage. Then, the efficiency of the proposed structure can be expressed as η=kη 1 +(1−k)η 1 η 2 &gt;η 1 η 2 . Obviously the overall efficiency can be improved by minimizing the power process times. In addition, reduced current stress also brings about higher efficiency due to reduced turn off losses;   6. Low turn off spikes. The snubber capacitor C 1  can effectively suppress the turn off spikes of the main switch, and in each switching cycle, its stored energy can also be released to the output through the coupling winding at the moment of main switch being turn on; and,   7. High power application. Two flyback transformers configuration has the potential to increase the power conversion rating and also release the thermal design difficulties due to distributed heat dissipation.       

   While the invention has been described, disclosed, illustrated and shown in various terms of certain embodiments or modifications which it has presumed in practice, the scope of the invention is not intended to be, nor should it be deemed to be, limited thereby and such other modifications or embodiments as may be suggested by the teachings herein are particularly reserved especially as they fall within the breadth and scope of the claims here appended.