Abstract:
A switching power converter and method of controlling an output voltage thereof using predictive sensing of magnetic flux provides a low-cost switching power converter via primary-side control using a primary-side winding. An integrator generates a voltage representing magnetic flux within a power magnetic element by integrating a voltage on a primary-side winding of the power magnetic element. A detection circuit detects the end of a half-cycle of post-conduction resonance in the power magnetic element after the energy level in the power magnetic element reaches zero. The voltage of the integrator is stored at the end of the post-conduction resonance half-cycle and is used to determine a sampling point prior to or equal to the start of post-conduction resonance in a subsequent switching cycle of the power converter. The primary-side winding voltage is then sampled at the sampling point, providing an indication of the output voltage of the power converter.

Description:
CROSS-REFERENCE TO RELATED APPLICATION  
       [0001]    This application is related to U.S. provisional application Ser. No. 60/482,580, filed Jun. 25, 2003 and from which it claims benefits under 35 U.S.C. §119(e). 
     
    
     
       BACKGROUND OF THE INVENTION  
         [0002]    1. Field of the Invention  
           [0003]    The present invention relates generally to power supplies, and more specifically to a method and apparatus for controlling a switching power converter entirely from the primary side of the power converter by predictive sensing of magnetic flux in a magnetic element.  
           [0004]    2. Background of the Invention  
           [0005]    Electronic devices typically incorporate low voltage DC power supplies to operate internal circuitry by providing a constant output voltage from a wide variety of input sources. Switching power converters are in common use to provide a voltage regulated source of power, from battery, AC line and other sources such as automotive power systems.  
           [0006]    Power converters operating from an AC line source (offline converters) typically require isolation between input and output in order to provide for the safety of users of electronic equipment in which the power supply is included or to which the power supply is connected. Transformer-coupled switching power converters are typically employed for this function. Regulation in a transformer-coupled power converter is typically provided by an isolated feedback path that couples a sensed representation of an output voltage from the output of the power converter to the primary side, where an input voltage (rectified line voltage for AC offline converters) is typically switched through a primary-side transformer winding by a pulse-width-modulator (PWM) controlled switch. The duty ratio of the switch is controlled in conformity with the sensed output voltage, providing regulation of the power converter output.  
           [0007]    The isolated feedback signal provided from the secondary side of an offline converter is typically provided by an optoisolator or other circuit such as a signal transformer and chopper circuit. The feedback circuit typically raises the cost and size of a power converter significantly and also lowers reliability and long-term stability, as optocouplers change characteristics with age.  
           [0008]    An alternative feedback circuit is used in flyback power converters in accordance with an embodiment of the present invention. A sense winding in the power transformer provides an indication of the secondary winding voltage during conduction of the secondary side rectifier, which is ideally equal to the forward drop of the rectifier added to the output voltage of the power converter. The voltage at the sense winding is equal to the secondary winding voltage multiplied by the turns ratio between the sense winding and the secondary winding. A primary power winding may be used as a sense winding, but due to the high voltages typically present at the power winding, deriving a feedback signal from the primary winding may raise the cost and complexity of the feedback circuit. An additional low voltage auxiliary winding that may also be used to provide power for the control and feedback circuits may therefore be employed. The above-described technique is known as “magnetic flux sensing” because the voltage present at the sense winding is generated by the magnetic flux linkage between the secondary winding and the sense winding.  
           [0009]    Magnetic flux sensing lowers the cost of a power supply by reducing the number of components required, while still providing isolation between the secondary and primary sides of the converter. However, parasitic phenomena typically associated with magnetically coupled circuits cause error in the feedback signal that degrade voltage regulation performance. The above-mentioned parasitics include the DC resistance of windings and switching elements, equivalent series resistance (ESR) of filter capacitors, leakage inductance and non-linearity of the power transformer and the output rectifier.  
           [0010]    Solutions have been provided in the prior art that reduce the effect of some of the above-listed parasitics. For example, adding coupled inductors in series with the windings or a leakage-spike blanking technique reduce the effect of leakage inductance in flyback voltage regulators. Other techniques such as adding dependence on the peak primary current (sensed switch current) to cancel the effect of the output load on sensed output voltage have been used. However, the on-resistance of switches typically vary greatly from device to device and over temperature and the winding resistances of both the primary and secondary winding also vary greatly over temperature. The equivalent series resistance (ESR) of the power converter output capacitors also varies greatly over temperature. All of the above parasitic phenomena reduce the accuracy of the above-described compensation scheme.  
           [0011]    In a discontinuous conduction mode (DCM) flyback power converter, in which magnetic energy storage in the transformer is fully depleted every switching cycle, accuracy of magnetic flux sensing can be greatly improved by sensing the voltage at a constant small value of magnetization current while the secondary rectifier is still conducting. However, no prior art solution exists that provides a reliable and universal method that adapts to the values of the above-mentioned parasitic phenomena in order to accurately sense the voltage at the above-mentioned small constant magnetization current point in DCM power converters.  
           [0012]    Therefore, it would be desirable to provide a method and apparatus for controlling a power converter output entirely from the primary, so that isolation bridging is not required and having improved immunity from the effects of parasitic phenomena on the accuracy of the power converter output.  
         SUMMARY OF THE INVENTION  
         [0013]    The above objective of controlling a switching power converter output entirely from the primary side with improved immunity from parasitic phenomena is achieved in a switching power converter apparatus and method. The power converter includes an integrator that generate a voltage corresponding to magnetic flux within a power magnetic element of the power converter. The integrator is coupled to a winding of the power magnetic element and integrates the voltage of the winding. A detection circuit detects an end of a half-cycle of post-conduction resonance that occurs in the power magnetic element subsequent to the energy level in the power magnetic falling to zero. The voltage of the integrator is stored at the end of a first post-conduction resonance half-cycle and is used to determine a sampling time prior to or equal to the start of a post-conduction resonance in a subsequent switching cycle of the power converter. At the sampling time, the auxiliary winding voltage is sampled and used to control a switch that energizes the power magnetic element.  
           [0014]    The foregoing and other objectives, features, and advantages of the invention will be apparent from the following, more particular, description of the preferred embodiment of the invention, as illustrated in the accompanying drawings, wherein like reference numerals indicate like components throughout.  
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0015]    [0015]FIG. 1 is a schematic diagram of a power converter in accordance with an embodiment of the present invention.  
         [0016]    [0016]FIG. 1B is a schematic diagram of a power converter in accordance with an alternative embodiment of the present invention.  
         [0017]    [0017]FIG. 2 is a waveform diagram depicting signals within the power converters of FIGS. 1 and 1B.  
         [0018]    [0018]FIG. 3 is a schematic diagram of a power converter in accordance with another embodiment of the present invention.  
         [0019]    [0019]FIG. 4 is a schematic diagram of a power converter in accordance with yet another embodiment of the present invention.  
         [0020]    [0020]FIG. 5 is a waveform diagram depicting signals within the power converters of FIGS. 3 and 4.  
         [0021]    [0021]FIG. 6 is a schematic diagram of a power converter in accordance with yet another embodiment of the present invention.  
         [0022]    [0022]FIG. 7 is a schematic diagram depicting details of an ESR-compensated control circuit in accordance with an embodiment of the present invention.  
         [0023]    [0023]FIG. 8 is a schematic diagram depicting details of an ESR-compensated control circuit in accordance with another embodiment of the present invention.  
     
    
     DETAILED DESCRIPTION OF THE EMBODIMENTS  
       [0024]    The present invention provides novel circuits and methods for controlling a power supply output voltage using predictive sensing of magnetic flux. As a result, the line and load regulation of a switching power converter can be improved by incorporating one or more aspects of the present invention. The present invention includes, alone or in combination, a unique sampling error amplifier with zero magnetization detection circuitry and unique pulse width modulator control circuits.  
         [0025]    [0025]FIG. 1 shows a simplified block diagram of a first embodiment of the present invention. The switching configuration shown is a flyback converter topology. It includes a transformer  101  with a primary winding  141 , a secondary winding  142 , an auxiliary winding  103 , a secondary rectifier  107  and a smoothing capacitor  108 . A resistor  109  represents an output load of the flyback converter. A capacitor  146  represents total parasitic capacitance present at an input terminal of primary winding  141 , including the output capacitance of the switch  102 , inter-winding capacitance of the transformer  101  and other parasitics. Capacitance may be added in the form of additional discrete capacitors if needed in particular implementations for lowering the frequency of the post-conduction resonance condition. The power converter of FIG. 3 also includes an input terminal  147 , a supply voltage terminal  143  which is a voltage derived from auxiliary winding  103  by means of a rectifier  113  and a smoothing capacitor  112 , a feedback terminal  144 , and a ground terminal  145 . Voltage VIN at the input terminal  147  is an unregulated or poorly regulated DC voltage, such as one generated by the input rectifier circuitry of an offline power supply. The power converter also includes a power switch  102  for switching current through the primary winding  141  from input terminal  147  to ground terminal  145 , a sample-and-hold circuit  124  connected to feedback terminal  144  via a resistive voltage divider formed by resistors  110  and  111 , an error amplifier circuit  123  having one of a pair of differential inputs connected to an output of sample-and-hold circuit  124  and having another differential input connected to a reference voltage REF, a pulse width modulator circuit  105  that generates a pulsed signal having a duty ratio as a function of an output signal of error amplifier circuit  123 , a gate driver  106  for controlling on and off states of power switch  102  in accordance with the output of the pulse width modulator circuit  105 , an integrator circuit  128  having an input connected to feedback terminal  144  and a reset input, a differentiator circuit  127  having an input connected to feedback terminal  144 , a zero-derivative detect comparator  126  having a small hysteresis and having one of a pair or differential inputs connected to the output of differentiator circuit  127 , and another differential input connected to an offset voltage source  131 , a blanking circuit  134  for selectively blanking the zero-derivative detect comparator  126  output, a sample-and-hold circuit  129  controlled by the output signal of the comparator  126  via the blanking circuit  134  for selective sampling-and-holding the output signal of the integrator circuit  128 ; a comparator  125  having one of a pair of differential inputs connected to the output of sample-and-hold circuit  129  and offset by a voltage source  130 , and another differential input connected to the output of integrator circuit  128 . The output of comparator  125  controls the sample-and-hold circuit  124 .  
         [0026]    Referring now to FIG. 1B, a forward power converter in accordance with an alternative embodiment of the present invention is depicted. Rather than auxiliary winding  103  being provided as a transformer winding, in the present embodiment, the feedback signal is provided by auxiliary winding  103  of an output filter inductor  145 . A free-wheeling diode  199  is added to the circuit to return energy from a power winding  198  of output filter inductor  145 , to capacitor  108  and load  109 . When switch  102  is enabled, a secondary voltage of positive polarity appears across winding  142  equal to input voltage VIN divided by turn ratio between windings  141  and  142 . Diode  107  conducts, coupling the power winding of inductor  198  between winding  142  and filter capacitor  108 . Energy is thereby stored in inductor  198 . When switch  102  is disabled, diode  107  becomes reverse biased, and diode  199  conducts, returning energy stored in inductor  198  to output filter capacitor  108  and load  109 . When the magnetic energy stored in inductor  198  fully depleted, inductor  198  enters post-conduction resonance (similar to that of transformer  101  in the circuit of FIG. 1). Therefore, auxiliary winding  103  provides similar waveforms as the circuit of FIG. 1 and provides a similar voltage feedback signal that are used by the control circuit of the present invention.  
         [0027]    Operation of the circuits of FIGS. 1 and 1B is depicted in the waveform diagram of FIG. 2, respecting the difference that auxiliary winding  103  of FIG. 1B is provided on output filter inductor  198 . Referring additionally to FIG. 2, at time Ton, power switch  102  is turned on. During the period of time between Ton and Toff, a linear increase of the magnetization current in primary winding  141  of flyback transformer  101  occurs. A voltage  201  of negative polarity and proportional to the input voltage VIN as determined by the turns ratio between auxiliary winding  103  and primary winding  141  will appear at feedback terminal  144 . (In the circuit of FIG. 1B, the feedback voltage is proportional to the difference between VIN divided by the turn ratio between windings  141  and  142  and the output voltage across capacitor  108 .) The feedback terminal  144  voltage causes a linear increase in the output voltage  202  of integrator  128 . The duration of the on-time of the power switch  102  is determined by the magnitude of the error signal at the output of error amplifier  123 .  
         [0028]    At time Toff, power switch  102  is turned off, interrupting the magnetization current path of primary winding  141  (or the power winding of inductor  198  in the circuit of FIG. 1B). Secondary rectifier  107  (or diode  199  in the circuit of FIG. 1B) then becomes forward biased and conducts the magnetization current of secondary winding  142  (or the power winding of inductor  198  in the circuit of FIG. 1B) to output smoothing capacitor  108  and load  109 . The magnetization current decreases linearly as the flyback transformer  101  (or inductor  198  in the circuit of FIG. 1B) transfers energy to output capacitor  108  and load  109 . A positive voltage  201  is then present at feedback terminal  144  (and similarly for the circuit of FIG. 1B after diode  107  ceases conduction and diode  199  conducts), having a voltage proportional to the sum of the output voltage across capacitor  108  and the forward voltage of rectifier  107  (or diode  199  in the circuit of FIG. 1B) and the proportion is determined by the turn ratio between auxiliary winding  103  and secondary winding  142  (or power winding  198  in the circuit of FIG. 1B). The feedback terminal  144  voltage causes the output voltage of integrator  128  to decrease linearly until, at time To, transformer  101  (or output filter inductor  198  in the circuit of FIG. 1B) is fully de-energized. At time To, rectifier  107  (or diode  199  in the circuit of FIG. 1B) becomes reverse biased, and the voltage across the windings of the transformer  101  (or inductor  198  in the circuit of FIG. 1B) reflects a post-conduction resonance condition as shown.  
         [0029]    The period of the post-conduction resonance is a function of the inductance of primary winding  141  and parasitic capacitance  146  (or the parasitic capacitance as reflected at the power winding of filter inductor  198  in the circuit of FIG. 1B). Differentiator circuit  127  continuously generates an output corresponding to the derivative of voltage  201  at feedback terminal  144 . The output of differentiator  127  is compared to a small reference voltage  131  by comparator  126 , in order to detect a zero-derivative condition at feedback terminal  144 . Comparator  126  provides a hysteresis to eliminate its false tripping due to noise at the feedback terminal  144 . Output voltage  202  of integrator  128  is sampled at time T2, when comparator  126  detects the zero-derivative condition at feedback terminal  144  (positive edge of comparator  126  output  204 ). Blanking circuit  134  disables the output of comparator  126 , only enabling sample-and-hold circuit  129  during post-conduction resonance. The blanking signal is represented by a waveform  205  and the output of blanking circuit  134  is represented by a waveform  206 .  
         [0030]    There are numerous ways to generate blanking waveform  205 . In the illustrative example, sampling is enabled at time T1 when the voltage at the feedback terminal  144  reaches substantially zero. The voltage at the output of sample-and-hold circuit  129  is offset by a small voltage  130  (AV of FIG. 2). During the next switching cycle, the previously sampled (held) voltage is compared to the output voltage of integrator  128  by comparator  125 . Comparator  125  triggers sample-and-hold circuit  124 , which samples the feedback voltage at the output of the resistive divider formed by resistors  110 ,  111  at time Tfb. Waveform  207  shows the timing of feedback voltage sampling by sample-and-hold circuit  124 . The sampled feedback voltage is compared to reference voltage REF by error amplifier  123 , which outputs an error signal that controls pulse width modulator circuit  105 .  
         [0031]    Every switching cycle, the output of integrator  128  is reset to a constant voltage level Vreset by a reset pulse  203  in order to remove integration errors. It is convenient to reset integrator  128  following time T2. However, in general, integrator  128  can be reset at any time with the exceptions of times Tfb and T1 which are sampling times.  
         [0032]    Since flyback transformer  101  (and inductor  198  in the circuit of FIG. 1B) is fully de-energized every switching cycle, the output of integrator  128  represents a voltage analog of the magnetization current in the transformer  101  (and magnetization current of filter inductor  198  in the circuit of FIG. 1B). Time To corresponds a point of zero magnetization current. Voltage offset AV sets a constant small from the actual secondary winding  142  zero-current point, and this a small offset in sampling time Tfb, at which the voltage at feedback terminal  144  is sampled. The technique described above eliminates the effect of most of the parasitic elements of the power supply, and substantial improvement of regulation of output voltage of the switching power converter is achieved.  
         [0033]    A method and apparatus in accordance with an alternative embodiment of the present invention are included in traditional peak current mode controlled pulse width modulator circuit to form a circuit as depicted in FIG. 3, wherein like reference designators are used to indicate like elements between the circuit of FIGS. 1 and 3. Only differences between the circuits of FIGS. 1 and 3 will be described below.  
         [0034]    Referring to FIG. 3, since the output voltage of the integrator  128  is a representation of the magnetic flux in transformer  101 , integrator  128  output is an indication of current conducted through power switch  102 . Pulse width modulator circuit includes a pulse width modulator comparator  132  and a latch circuit  133 . In operation, when the output voltage of integrator  128  the output voltage of error amplifier  123 , comparator  132  resets latch  133  and turns off power switch  102 . Latch  133  is set with a fixed frequency Clock signal at the beginning of the next switching cycle, initiating the next turn-on of the switch  102 .  
         [0035]    [0035]FIG. 4 depicts a switching power converter in accordance with yet another embodiment of the present invention that is similar to the circuit of FIG. 3, but is set up to operate in critically discontinuous (boundary) conduction mode of flyback transformer  101 . Unlike the power converter of FIG. 3, which operates at a constant switching frequency determined by the frequency of the Clock signal, the circuit of FIG. 4 is free running. A free running operating mode is provided by connecting the output of blanking circuit  134  to the “S” (set) input of latch  133 . Operation of the circuit of FIG. 4 is illustrated in the waveform diagrams of FIG. 5. Referring to FIGS. 6 and 7, waveform  301  represents the voltage at feedback terminal  144 , waveform  302  shows the output voltage of the integrator circuit, and waveform  303  shows the Reset timing of the integrator  128 . The output of zero-derivative detect comparator  126  is depicted by waveform  304 . Waveforms  305 ,  306  and  307  show the blanking  134 , the integrator sample-and-hold  129  and feedback sample-and-hold  124  timings, respectively. Operation of the power converter circuit of FIG. 4 is similar to the one of FIG. 3, except that latch circuit  133  is reset by the output of blanking circuit  134 . The reset occurs when comparator  126  detects a zero-derivative condition in feedback terminal  144  output voltage  301  during post-conduction resonance. Therefore, power switch  102  is turned on after one half period of the post conduction resonance at the lowest possible voltage across switch  102 . The above-described “valley” switching technique minimizes power losses in switch  102  due to discharging of parasitic capacitance  146 . At the same time, the transformer  101  is operated in the boundary conduction mode, since the next switching cycle always starts immediately after the entire magnetization energy is transferred to the power supply output. Operating the transformer  101  in the critically discontinuous conduction mode reduces power loss and improves the efficiency of the switching power converter of FIG. 4.  
         [0036]    Indirect current sensing by synthesizing a voltage corresponding to magnetization current (as performed in the control circuits of FIGS. 3, 4 and  6 ) enables construction of single stage power factor corrected (SS-PFC) switching power converters. One example of such an SS-PFC switching power converter is shown in FIG. 6. The control circuit is identical to that of FIG. 4, only the switching and input circuits differ. Common reference designators are used in FIGS. 4 and 6 and only differences will be described below.  
         [0037]    The power converter of FIG. 6 includes a power transformer  101  with two primary windings  141 , two bulk energy storage capacitors  135  with a series connected diode  190 , in addition to all other elements of the power converter of FIG. 4. The input voltage VIN is a full wave rectified input AC line voltage. In operation, referring to FIGS. 5 and 6, when power switch  102  is turned on at time Ton, the voltage VIN is applied across a boost inductor  136  via a diode  137 , causing a linear increase in the current through inductor  136 . At the same time, a substantially constant voltage from bulk energy storage capacitors  135  is applied across primary windings  141  causing transformer  101  to store magnetization energy. Diode  190  is reversed-biased during this period. Between times Ton and Toff, power switch  102  conducts a superposition of magnetization currents of the transformer  101  and boost inductor  136 . Following time Toff, transformer  101  transfers its stored energy via diode  107  to capacitor  108  and load  109 . Simultaneously, boost inductor  136  transfers its energy to bulk energy storage capacitors  135  via primary windings  141  and forward biased diode  190 .  
         [0038]    Boost inductor  136  is designed to operate in discontinuous conduction mode. Therefore, its magnetization current is proportional to the input voltage VIN, inherently providing good power factor performance, as the average input impedance has little or no reactive component. Diode  137  ensures discontinuous conduction of boost inductor  136  by blocking reverse current. A peak current mode control scheme that maintains peak current in power switch  102  in proportion to the output of voltage error amplifier  123 , is not generally desirable in the power converter of FIG. 6. Since the current through power switch  102  is a superposition of the currents in boost inductor winding  136  and transformer primary windings  141 , keeping the power switch current proportional to the voltage error signal tends to distort the input current waveform.  
         [0039]    In summary, with respect to the control circuit of FIG. 6, the voltage error signal is made independent of the current in boost inductor  136 , while the voltage error signal set proportional to the magnetization current in the transformer  101 . Therefore, the switching power converter of FIG. 6 inherently provides good power factor performance. In addition, the above-described control circuit eliminates the need for direct current sensing. The method of the control circuit described above also provides an inherent output over-current protection when the voltage error signal is limited.  
         [0040]    While the switching power converters of FIGS. 4 and 6 eliminate the effect of most of the parasitics in a power converter, a small error in the output voltage regulation is still present due to series resistance (ESR) of output capacitor  108 . The current into the capacitor  108  is equal to (I2−Io) where  12  is current in secondary winding  142 , and Io is the output current of the switching power converter. The output voltage deviation from the average output voltage can be expressed as ESR*(I2−Io), where ESR is equivalent series resistance of capacitor  108 . The sampling error is represented by the deviation from the average output voltage at a time when  12  is zero. Therefore, the above-described error is equal to (−Io*ESR). FIG. 7 depicts a compensation resistor  138  connected between the output of voltage error amplifier  123  and the output of the resistive divider formed by resistors  110 ,  111 , which can be added to the switching power converters of FIGS. 4 and 6 to cancel the above-described regulation error, since the voltage at the output of error amplifier  123  is representative of the power converter output current Io.  
         [0041]    The circuit of FIG. 7 compensates for output voltage error due to ESR of capacitor  108  for a given duty ratio of power switch  102 . The value of resistor  138  is selected in inverse proportion to (1-D), where D is the duty ratio of the power switch  102 . When more accurate compensation is needed, a circuit as depicted in FIG. 8 may be implemented. The circuit of FIG. 8 includes a compensation resistor  138 , a low pass filter  139  and a chopper circuit  140 . In operation, chopper circuit  140  corrects the compensation current of resistor  138  by factor of (1-D), chopping the output voltage of error amplifier  123  using the inverting output signal of the pulse width modulator latch  133 . The switching component of the compensation signal is filtered using low pass filter  139 .  
         [0042]    The present invention introduces a new method and apparatus for controlling output voltage of magnetically coupled isolated switching power converters that eliminate a requirement for opto-feedback, current sense resistors and/or separate feedback transformers by selective sensing of magnetic flux. Further, the present invention provides high switching power converter efficiency by minimizing switching losses. The present invention is particularly useful in single-stage single-switch power factor corrected AC/DC converters due to the indirect current sensing technique of the present invention, but may be applied to other applications where the advantages of the present invention are desirable. While the illustrative examples include an auxiliary winding of a power transformer or output filter inductor for detecting magnetic flux and thereby determining a level of magnetic energy storage, the circuits depicted and claimed herein can alternatively derive their flux measurement from any winding of a power transformer or output filter inductor. Further, the measurement techniques may be applied to non-coupled designs where it may be desirable to detect the flux in an inductor that is discontinuously switched between an energizing state and a load transfer state.  
         [0043]    While the invention has been particularly shown and described with reference to the preferred embodiments thereof, it will be understood by those skilled in the art that the foregoing and other changes in form, and details may be made therein without departing from the spirit and scope of the invention.