Abstract:
A method and apparatus for estimation of frequency offset in wireless communications. Symbols to be transmitted are arranged in blocks of K symbols. The symbol blocks and their replicas are interleaved and then encoded into K sequences of coded symbols in space and time domains. The K coded symbol sequences are transmitted from K transmit antennas, while baseband signals carrying the coded symbols are received via J receive antennas and converted into J groups of intermediate signals. Thereafter, an interlaced operation and coherent combination is performed on each of the J intermediate signal groups. J sets of computed signals are thus generated and then combined in the space and time domains to develop a set of combined signals. Consequently, a frequency offset is estimated on the basis of this combined signal set.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    The invention relates to wireless communication, and more particularly to a transmit diversity scheme used in a coherent system for estimation of frequency offset in a flat-fading environment.  
           [0003]    2. Description of the Related Art  
           [0004]    In the past two decades, wireless communication has grown with unprecedented speed from early radio paging, cordless telephone, and cellular telephony to today&#39;s personal communication and computing devices. These commercial wireless applications have had a profound impact on today&#39;s business world and people&#39;s daily lives. It is likely that future breakthroughs in wireless communication will be driven largely by high data rate applications. Increasing the link or channel bandwidth is a simple but costly remedy. It has been shown that Multiple-Input Multiple-Output (MIMO) systems are a viable approach to provide significant capacity improvement over conventional wireless antenna systems. MIMO links may play an important role in next-generation wireless systems.  
           [0005]    The system performance of current wireless communication systems is limited by the major channel impairment, that is, signal fading generally caused by multipath propagation. Theoretically, the most effective technique to mitigate multipath fading in a wireless channel is transmitter power control. However, there are two fundamental problems with this approach. The major problem is the required transmitter dynamic range. The second problem is that the transmitter does not have any knowledge of the channel experienced by the receiver. Hence, the channel information has to be fed back from the receiver to the transmitter, which results in throughput degradation and considerable added complexity to both the transmitter and the receiver.  
           [0006]    Other effective techniques such as space, time, and frequency diversity can be used to mitigate the effect of fading. Frequency and time diversity can be exploited from time-varying multipath channels while space (or antenna) diversity is achieved using multiple antennas. In the classical space diversity scheme, multiple antennas are often employed to provide diversity at the receiver. Such receiver diversity approaches may not be desirable for mobile handsets because of the cost, size and power considerations. Therefore, transmit diversity schemes have become one of the most interesting topics in wireless communications. U.S. Pat. No. 6,185,258, issued to Alamouti et al., disclose a transmitter diversity technique for wireless communications. The technique provides good diversity gain, but is very sensitive to channel estimation errors. In this patent, the receiver can only be applicable to a path fading, not channel fading, environment since it cannot deal with the carrier frequency offset. In reality, the frequency offset always exists due to the difference of the oscillator frequencies between the transmitter and the receiver or the Doppler frequency shift effect for mobile communications. Tarokh et al. disclose a differential modulation for two transmit antennas and one receive antenna based on Alamouti&#39;s method, see “A differential detection scheme for transmit diversity,”  IEEE J. Select. Areas Commun ., Vol. 18, No. 7 pp. 1169-1194, July, 2000. In this scheme, the receiver executes differential detection and requires no channel state information to accomplish low decoding complexity. Unfortunately, the differential encoding scheme without estimation of frequency offset is suited only to a noncoherent system but not to a coherent system.  
           [0007]    In view of the above, what is needed is a novel transmit diversity scheme applicable to a coherent system, unencumbered by the limitations associated with the prior art.  
         SUMMARY OF THE INVENTION  
         [0008]    It is an object of the present invention to provide a method used in a coherent system for effective estimation of frequency offset in a flat-fading environment.  
           [0009]    It is another object of the present invention to provide a mobile communication arrangement suitable for integrated circuits, which offers the benefits of space and time diversity without the use of channel estimation.  
           [0010]    According to one aspect of the invention, an apparatus for wireless transmission of symbols via multiple antennas includes a block replicater, a space-time encoder and an output stage. The block replicater arranges symbols to be transmitted in blocks of K symbols, where K is the number of transmit antennas. The block replicater subsequently outputs each symbol block followed by a replica thereof. The space-time encoder generates K sequences of coded symbols by respectively encoding each symbol block and its replica in space and time domains. Then, the output stage transmits the K coded symbol sequences from K antennas, respectively.  
           [0011]    According to another aspect of the invention, an apparatus for wireless reception of signals via multiple antennas is disclosed. The inventive apparatus includes a space-time combiner and an estimator, J distributors and J block processors, where J is the number of receive antennas. The J distributors receive respective baseband signals from corresponding antennas and each distributor converts the received baseband signal into a group of intermediate signals. Each of the J block processors performs an interlaced operation and coherent combination on the corresponding group of intermediate signals to generate a set of computed signals. The space-time combiner then combines J sets of computed signals from the J block processors in space and time domains, thereby generating a set of combined signals. Thus, the estimator can estimate a frequency offset based on the set of combined signals.  
           [0012]    According to yet another aspect of the invention, a method for estimation of frequency offset in wireless communications is provided. The first step of the method comprises arrangement of symbols to be transmitted in blocks of K symbols as well as interleaving of each symbol block and its replica, where K is the number of transmit antennas. Thereafter, each symbol block and its replica are encoded in space and time domains into K sequences of coded symbols. The K coded symbol sequences are then transmitted from K transmit antennas, separately. Hence, baseband signals including the coded symbols are separately received via J receive antennas, where J is the number of receive antennas. The received baseband signals are converted into J groups of intermediate signals, respectively. After that, an interlaced operation and coherent combination is performed on each of the J intermediate signal groups to generate J sets of computed signals respectively. The J computed signal sets are combined together in the space and time domains to derive a set of combined signals. As a result, a frequency offset is estimated on the basis of the combined signal set. 
       
    
    
     DESCRIPTION OF THE DRAWINGS  
       [0013]    The present invention will be described by way of exemplary embodiments, but not limitations, illustrated in the accompanying drawings in which like references denote similar elements, and in which:  
         [0014]    [0014]FIG. 1 is a block diagram illustrating a transmitter according to the invention;  
         [0015]    [0015]FIG. 2 is a block diagram illustrating a receiver according to the invention;  
         [0016]    [0016]FIG. 3 is a detailed block diagram illustrating a block processor of FIG. 2;  
         [0017]    [0017]FIG. 4 is a detailed block diagram illustrating a space-time combiner of FIG. 2;  
         [0018]    [0018]FIG. 5 is a detailed block diagram illustrating an estimator of FIG. 2;  
         [0019]    FIGS.  6 ˜ 8  are graphs showing simulation results of the invention. 
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0020]    [0020]FIG. 1 shows a block diagram focused on a transmitter in accordance with an embodiment of the invention. The transmitter  100  includes multiple antennas to provide space diversity, although, for illustrative simplicity, the illustrative example employs only two transmit antennas TX 1  and TX 2 . Note that binary baseband data to be transmitted is digitally modulated with binary phase shift keying (BPSK) or quadrature phase shift keying (QPSK). In aid of frequency offset estimation, a symbol stream {S 1 ,S 2 , . . . , S i ,S j , . . . } is first fed to a block replicater  110  of the transmitter  100 . The block replicater  110  arranges the symbols to be transmitted in blocks of K symbols, where K is the number of transmit antennas. Since K equals 2 in the exemplary embodiment of FIG. 1, the block replicater  110  groups the transmitted symbols into blocks of two adjacent symbols {S i ,S j }. Thereafter, the block replicater  110  sends every two adjacent symbols {S i ,S j } to a space-time encoder  120  and sends the two symbols {S i ,S j } once again. In other words, the block replicater  110  interleaves each symbol block {S i ,S j } and its replica as {S i ,S j ,S i ,S j } and outputs each symbol block followed by a replica thereof. The output of the block replicater  110 , for instance, is {S 1 ,S 2 ,S 1 S 2 , . . . ,S i ,S j ,S i ,S j , . . . }.  
         [0021]    The space-time encoder  120  generates K sequences of coded symbols by respectively encoding each symbol block and its replica in space and time domains. To provide space and time diversity, a space-time block encoding is performed on each symbol block and its replica. In the two-antenna embodiment, the sequence {S 1 ,S 2 ,S 1 ,S 2 , . . . ,S i ,S j ,S i ,S j , . . . } is encoded into two coded symbol sequences, i.e., a first sequence of coded symbols {S 1 ,−S 2   * ,S 1 ,−S 2   * , . . . ,S i ,−S j   * ,S i ,−S j   * , . . . } and a second sequence of coded symbols {S 2 ,S 1   * ,S 2 ,S 1   * , . . . ,S j ,S i   * ,S j ,S i   * , . . . }, where * appended to a symbol represents the complex conjugate of that symbol. It should be appreciated that the principles of the invention are applicable to transmitters with more than two antennas. The coded symbol sequences are sent to an output stage  130  where upconversion and power amplification are performed. Then, the output stage  130  applies the first coded symbol sequence { . . . ,S i ,−S j   * ,S i ,−S j   * , . . . } to the antenna TX 1 , and the second coded symbol sequence { . . . ,S j ,S i   * ,S j ,S i   * , . . . } to the antenna TX 2 , and transmits them from the two antennas, respectively. At each time slot, two symbols are simultaneously transmitted from the antennas TX 1  and TX 2 . Table 1 illustrates the encoding and transmission sequence for the transmitter  100  in accordance with the invention.  
                                                                                                             TABLE 1                           The Order of Coded Symbols                Time                t 0     t 1     t 2     t 3     t 4     t 5     t 6     t 7     . . .                        Antenna TX 1     S 1     −S 2 *   S 1     −S 2 *   S 3     −S 4 *   S 3     −S 4 *   . . .       Antenna TX 2     S 2      S 1 *   S 2      S 1 *   S 4      S 3 *   S 4      S 3 *   . . .                  
 
         [0022]    Still referring to FIG. 1, in this case, a diversity order of 2J with two transmit antennas and J receive antennas is provided. In a flat-fading environment, channels between the transmit antennas TX 1 , TX 2  and the jth receive antenna RX j  are denoted by h 1,j  and h 2,j , respectively, assuming that fading is constant within two consecutive block durations. Table 2 here summarizes the definition of channels between the transmit and receive antennas.  
                                                                         TABLE 2                                   Antenna   Antenna       Antenna       Antenna           RX 1     RX 2     . . .   RX j     . . .   RX J                                      Antenna TX 1     h 1,1     h 1,2     . . .   h 1,j         h 1,J         Antenna TX 2     h 2,1     h 2,2     . . .   h 2,j     . . .   h 2,J                    
 
         [0023]    [0023]FIG. 2 shows a block diagram of a receiver in accordance with the invention. Baseband signals x (1) (n) through x (J) (n) received via the antennas RX 1  through RX J  are downconverted by J input stages  210  of the receiver  200 , where J is the number of receive antennas. The baseband signals x (1) (n)˜x (J) (n) are then fed to J distributors  220 , respectively. At a given time interval m, the jth distributor  220  converts the received baseband signal x (j) (n) into a group of intermediate signals:  
         [0024]    y i   (j) (m)=x (j) (4(m−1)+i), for i=1, 2, 3, 4 and jεJ  
         [0025]    where n=4m. Furthermore, the jth group of intermediate signals y (j) (m) at the given time interval m can be expressed as  
           y   1   (j) ( m )= x   (j) (4 m− 3)=( S   2(m−1)+1   ·h   1,j   +S   2(m−1)+2   ·h   2,j )· e   j2π·Δf·t     0     +N   1   (1)  
           y   2   (j) ( m )= x   (j) (4 m− 2)=( −S   2(m−1)+2   *   ·h   1,j   +S   2(m−1)+1   *   ·h   2,j )· e   j2π·Δf·(t     0     +T)   +N   2   (2)  
           y   3   (j) ( m )= x   (j) (4 m− 1)=( S   2(m−1)+1   ·h   1,j   +S   2(m−1)+2   ·h   2,j )· e   j2π·Δf·(t     0     +2T)   +N   3   (3)  
           y   4   (j) ( m )= x   (j) (4 m )=( −S   2(m−1)+2   *   ·h   1,j   +S   2(m−1)+1   *   ·h   2,j )· e   j2π·Δf·(t     0     +3T)   +N   4   (4)  
         [0026]    where Δf is an actual frequency offset, t 0  is an initial time, T is a symbol period for every symbols S 2(m−1)+1  and S 2(m−1)+2 , and N 1  to N 4  represent additive white Gaussian noise.  
         [0027]    The J intermediate signal groups are sent to J block processors  230  where interlaced operations and coherent combinations are performed to generate J sets of computed signals r 1   (j) (m) and r 2   (j) (m) for j=1, . . . ,J. FIG. 3 is a more detailed block diagram of an embodiment of the block processor  230 . As depicted, each block processor is comprised of two conjugate operators  301  and  303 , two complex multipliers  311  and  313 , two adders  321  and  323 , a real part operator  331  and an imaginary part operator  333 . As depicted, y 2   (j) (m) and y 3   (j) (m) are sent to the conjugate operators  301 ,  303  where conjugations are employed to respectively obtain y 2   (j)* (m) and y 2   (j)* (m). Here * superscript defines the complex conjugate operation. To eliminate the influence of unknown channels and signals, y 2   (j)* (m) and y 3   (j)* (m) are interlaced and fed to the complex multipliers  311  and  313  where product A of y 1   (j) (m) and y 3   (j)* (m) and product B of y 2   (j)* (m) and y 4   (j) (m) are calculated. The sum C of the products A and B is added by the adder  321 . In a similar fashion, the sum D of the product A and a negative of the product B is added by the adder  323 . Finally, the real part operators  331  return the real part of the sum C and the imaginary part operator  333  return the imaginary part of the sum D, thereby forming the jth set of computed signals  
           r   1   (j) ( m )= Re[y   1   (j) ( m )· y   3   (j)* ( m )+ y   2   (j)* ( m )· y   4   (j) ( m )]  (5)  
           r   2   (j) ( m )= Im[y   1   (j) ( m )· y   3   (j)* ( m )− y   2   (j)* ( m )· y   4   (j) ( m )]  (6)  
         [0028]    where Re[ ] and Im[ ] represent the real and imaginary part of complex numbers, respectively. Substituting equations (1)˜(4) into (5) and (6) yields  
                 r   1     (   j   )            (   m   )       =         {       ∑     k   =   1     2                     |     h     k   ,   j            |   2         }     ·     {       ∑     k   =   1     2                     |     S       2        (     m   -   1     )       +   k            |   2         }     ·     cos        (     2        π   ·   Δ                     f   ·   2        T     )         +     N   1   ′               (   7   )                   r   2     (   j   )            (   m   )       =         -     {       ∑     k   =   1     2                     |     h     k   ,   j            |   2         }       ·     {       ∑     k   =   1     2                     |     S       2        (     m   -   1     )       +   k            |   2         }     ·     sin        (     2        π   ·   Δ                     f   ·   2        T     )         +     N   2   ′               (   8   )                               
 
         [0029]    where N 1 ′ and N 2 ′ represent the additive white Gaussian noise after performing the interlaced operation and coherent combination. From the equations (7) and (8), the unknown channel transfer functions h k,j  are coherently combined, as are the unknown signals S 2(m−1)+k . In addition, the signal-to-noise ratio (SNR) of the sinusoidal waveforms of r 1   (j) (m) and r 2   (j) (m) is increased due to the time diversity.  
         [0030]    After that, the space-time combiner  240  combines the J computed signal sets from the J block processors  230  in space and time domains. FIG. 4 shows a more detailed block diagram of an embodiment of the space-time combiner  240 . As depicted, the space-time combiner  240  is made up of two accumulators  401  and  403  for a J-dimensional space domain and two accumulators  411  and  413  for an M-dimensional time domain. With these accumulators, the space-time combiner  240  takes double summations of r 1   (j) (m) and r 2   (j) (m) over the J-dimensional space and M-dimensional time domains as follows:  
         z   1     =       ∑     m   =   1     M                       ∑     j   =   1     J                       r   1     (   j   )            (   m   )                     z   2     =       ∑     m   =   1     M                       ∑     j   =   1     J                       r   2     (   j   )            (   m   )                                 
 
         [0031]    where z 1  and z 2  form the set of combined signals. In this way, the space-time diversity gain is achieved, the SNR is further increased, and the frequency offset estimation can thus be improved.  
         [0032]    Turning now to FIG. 5, the estimator  250  includes an arctangent unit  501  and a multiplier  511 . Based on the set of combined signals z 1  and z 2 , the estimator  250  can attain an estimated frequency offset. In this regard, the arctangent unit  501  performs the trigonometric arctangent operation on −z 2 /z 1  to estimate a phase offset  
         Δ        φ   ^       =       tan     -   1            (       -     z   2         z   1       )                             
 
         [0033]    where Δ{circumflex over (φ)} is an estimated phase offset. Then, the estimated phase offset is fed to the multiplier  510  where Δ{circumflex over (φ)} is multiplied by a constant 1/(2π·2T). As a result, the estimated frequency offset is expressed as  
         Δ        f   ^       =       1     2        π   ·   2        T              tan     -   1            (       -     z   2         z   1       )                               
 
         [0034]    [0034]FIGS. 6 through 7 demonstrate simulation results of the invention. In the simulation, the actual frequency offset is assumed to be 30 KHz in a Rayleigh flat-fading environment. It is also assumed that there are two transmit antennas and J receive antennas (J=1, 2, 4). It is also assumed that the symbol period T is 5 μsec, so Δf·T is 0.15. As shown in FIG. 6, with one receive antenna (J=1) and M data samples (M=100, 300, 500), the root mean square error (RMSE) of the estimated frequency offset Δ{circumflex over (f)} decreases when SNR (E b /N 0 ) increases. Apart from E b /N 0 , FIG. 6 presents that the more data samples employed, the lower the RMSE of Δ{circumflex over (f)}. In FIG. 7, if J=1 and E b /N 0  equals 10 dB, 30 dB, and 50 dB, the RMSE of Δ{circumflex over (f)} declines rapidly if the number of data samples M is less than 300. When M reaches 1200, the RMSE of Δ{circumflex over (f)} converges toward a constant. In FIG. 8, if E b /N 0  is 10 dB and the number of data samples M is 100, 300 and 500, the RMSE of Δ{circumflex over (f)} is reduced as the receive antenna number grows.  
         [0035]    Accordingly, the invention provides a novel scheme to estimate the frequency offset of a flat-fading environment. The scheme of the invention offers the benefits of space and time diversity without the requirement of channel estimation. It is also shown that the invention is well-suited for MIMO wireless systems. Due to low computation complexity, the inventive arrangement is very simple to implement in an integrated circuit.  
         [0036]    While the invention has been described by way of example and in terms of the preferred embodiments, it is to be understood that the invention is not limited to the disclosed embodiments. To the contrary, it is intended to cover various modifications and similar arrangements (as would be apparent to those skilled in the art). Therefore, the scope of the appended claims should be accorded the broadest interpretation so as to encompass all such modifications and similar arrangements.