Abstract:
A method is disclosed for filtering a received RF signal modulated with a time series sampled data signal, each data sample occurring within a bit time. At least a portion of the signal is filtered such that in the frequency domain a portion of the high frequency energy therein is rejected to provide a filtered signal. In the time domain, substantial attenuation regions are disposed forward in time with the step of filtering such that the attenuation regions are disposed within the bit time of subsequent data samples of the time series sampled data signal in the time domain. Sampling of a given filtered received time series sampled data signal occurs substantially proximate in time to the substantial attenuation contributed by prior received data samples.

Description:
TECHNICAL FIELD OF THE INVENTION  
       [0001]     The present invention pertains in general to digital filters and, more particularly, to a digital filter that filters an incoming demodulated Offset-QPSK data stream in such a manner so as to reduce adjacent channel interference without introducing Inter-Symbol Interference (ISI).  
       BACKGROUND OF THE INVENTION  
       [0002]     Wireless transmission technologies have seen increased use due to the explosion of the wireless communication devices that allow computers to communicate with network interfaces, hands-free telephone handsets to communicate with a base station telephone, and other applications. In order to facilitate the transmission of data between one wireless transmitter and a wireless receiver, data is typically modulated onto a carrier with some type of modulation scheme and then the carrier transmitted to the receiver. The receiver then receives the carrier, demodulates the carrier and extracts the data therefrom as recovered data. Typical modulation schemes utilize a frequency shift key (FSK) modulation scheme or a phase shift key (PSK) modulation scheme. To obtain greater bandwidth efficiency, M-ary modulation schemes are the modulation scheme of choice. One such type of M-ary PSK utilizes quadrature modulation wherein in-phase and quadrature components of the signal are generated and, when the channels are independent of each other, this is known as quadrature PSK (QPSK). An even further variation of this, which is utilized for band-limited, non-linear channels, is offset quadrature phase shift keying (O-QPSK) and minimum shift keying (MSK). In a non-linear channel, the spectral side lobe of a filtered QPSK signal tends to be restored to its initial characteristics prior to filtering, whereas with O-QPSK and MSK, the signal envelope is constant, which makes these modulation techniques impervious to channel non-linear areas. The choice of O-QPSK (sometimes referred to as staggered QPSK), the I- and Q-bit streams are offset in time by one bit period, T c . These are well known techniques and the O-QPSK modulation scheme is usually found in applications that are associated with band-limited, non-linear channels wherein band-limiting is necessary to meet spectrum occupancy allocations.  
         [0003]     The power spectral density (PSD) of an O-QPSK modulated signal results in a number of lobes that, when processed through a modulator, and utilizing some type of matched filter, will result in recovery of all of the spectral information therefrom. However, in normal environments, there can be interference from adjacent channels that occurs within the energy spectrum of the O-QPSK recovered signal. For example, in a typical O-QPSK system, the main lobe in the energy spectrum has a width of +/−1.5 MHz. Thus, to reconstruct the signal and recover all the energy, the matched filter must recover the energy through the entire width of the main lobe and also the width of the smaller lobes that extend out from the +/−1.5 MHz by increments of 1.0 MHz. However, one interference source is what is referred to as “Blue Tooth” systems that will have a potential channel that is separated by 1.0 MHz from the center frequency of the O-QPSK signal. Therefore, there will be a potential source of interference that is disposed at a center frequency of 1.0 MHz from the center of the current channel with a band-width of +/−0.5 MHz. Therefore, at a distance of 0.5 MHz from the center frequency of the given channel, there is a potential for interference from the adjacent channel. Thus, it would be desirable to provide a band-pass filter that will filter out the adjacent channel. Although approximately 80% of the power spectral density will be recovered if the band pass filter cuts off sharply at +/−0.5 MHz, such filtering can result in a high degree of Inter-Symbol Interference (ISI). Thus, one has to make trade-offs between capturing the entire energy in the energy spectrum associated with the captured signal and thus being required to tolerate the adjacent channel interference, or filtering out the adjacent channel interference and then tolerating the ISI associated with that filtering function.  
       SUMMARY OF THE INVENTION  
       [0004]     The present invention disclosed and claimed herein, in one aspect thereof, comprises a method for filtering a received RF signal modulated with a time series sampled data signal, each data sample occurring within a bit time. At least a portion of the signal is filtered such that in the frequency domain a portion of the high frequency energy therein is rejected to provide a filtered signal. In the time domain, substantial attenuation regions are disposed forward in time with the step of filtering such that the attenuation regions are disposed within the bit time of subsequent data samples of the time series sampled data signal in the time domain. Sampling of a given filtered received time series sampled data signal occcurs substantially proximate in time to the substantial attenuation contributed by prior received data samples. BRIEF DESCRIPTION OF THE DRAWINGS  
         [0005]     For a more complete understanding of the present invention and the advantages thereof, reference is now made to the following description taken in conjunction with the accompanying Drawings in which:  
         [0006]      FIG. 1  illustrates an overall diagrammatic view of a transmission system with an interfering signal;  
         [0007]      FIG. 2  illustrates a diagrammatic view of a transmitter and receiver operating in accordance with the disclosed system;  
         [0008]      FIG. 3  illustrates a more detailed diagrammatic view of the receiver;  
         [0009]      FIG. 4  illustrates a diagram of the time domain of a half-sine symbol;  
         [0010]      FIG. 5  illustrates a frequency plot of the power spectral density of the O-QPSK signal;  
         [0011]      FIG. 6  illustrates a plot of the power spectral density for an O-QPSK signal with a raised cosine filter;  
         [0012]      FIG. 6   a  illustrates the channel response for the filtered O-QPSK signal;  
         [0013]      FIG. 7  illustrates the power spectral density for the unfiltered O-QPSK signal as a solid line and the power spectral density for the baseband filter response of one embodiment as a dotted line;  
         [0014]      FIG. 7   a  illustrates the power spectral density for the channel response after filtering;  
         [0015]      FIG. 8  illustrates a plot of one example of the O-QPSK filter;  
         [0016]      FIG. 9  illustrates the channel time response for the raised cosine filter of  FIG. 6   a  and the filter response of  FIG. 7   a;    
         [0017]      FIG. 10  illustrates the time response for multiple half-sine signals in a received signal stream;  
         [0018]      FIG. 11  illustrates a FIR filter implementation of the baseband filter of the disclosed embodiment; and  
         [0019]      FIG. 12  illustrates a diagrammatic view of the IF portion of the receiver.  
     
    
     DETAILED DESCRIPTION OF THE INVENTION  
       [0020]     Referring now to the drawings, and more particularly to  FIG. 1 , there is illustrated a diagrammatic view of a communications system. The primary communications system is comprised of a transmitter  102  and the receiver  104 . The transmitter  102  has a transmitting antennae  106  associated therewith for transmitting over an RF link  108  to a receiving antennae  110  on the receiver  104 . The transmitter  102  is operable to receive input data, modulate that input data onto a carrier and transmit that carrier over the communication link  108  to the receiver  104 . The receiver  104  is operable to receive the carrier, demodulate the carrier and then recover data therefrom and output this recovered data. However, the quality of the data recovery operation in the form of demodulation, etc., is a function of the quality of the communication link, the type of modulation and, also, the presence of noise in the system. Noise, as is well known, can come from many sources. One type of noise can actually come from other transmission systems. In the embodiment of  FIG. 1 , there is illustrated an interfering transmitter  120  which is operable to transmit a signal  122  into the transmission space of the receiver  104 . Thus, the receiver  104  can receive the transmission signal  122  on its antennae  110 . Typically, the receivers  104  have some type of band-pass or receive filter that will reject signals that are “out-of-band.” If the filter is narrow enough, only signals from the transmitter  102  will be received. However, depending upon the modulation, the filtering must recover energy from frequencies outside of the narrow band disposed about the carrier and bounded by the filter. Thus, depending upon the modulation technique, it is possible for some energy from the signal  122  from the transmitter  120  to affect the signal quality at the receiver  104  over the channel  108 . This is referred to as “adjacent channel interference.” As will be described hereinbelow, the receiver  104  incorporates a filtering technique that will reject adjacent channel interference and recover less than all of the energy in the received signal without substantially affecting the quality of the received signal.  
         [0021]     Referring now to  FIG. 2 , there is illustrated a diagrammatic view of the transmitter  102  and receiver  104 . In the disclosed embodiment, the type of modulation utilized is offset-quadrature phase shift key (O-QPSK) modulation. This typically requires I- and Q-channels offset in phase by 90° whereas each of the channels operates with half-sine signals representing the digital values and referred to as “symbols.” This will be described in more detail hereinbelow. The transmitter  102  receives binary data into an I- and Q-signal generator  202  to generate the I- and Q-signals. The I-signal is input to an up converter  204  that is driven by an oscillator  206  to basically modulate the I-signal onto a carrier that is provided by the oscillator  206 . The Q-signal is input to an up converter  208  which receives a 90° shifted signal from the oscillator  206  provided by a quadrature phase shift block  210 . The output of the up converter  204  and the up converter  208  are summed together in a summing block  212  to provide an output on the communication channel  108 .  
         [0022]     The receiver  104  receives the signal on an input node  220 . The receive signal is input to a down converter  222  which receives a clock signal from a clock recovery block  224 , which clock recovery block  224  recovers the clock from the receive signal on node  220 . This provides the I-signal at an intermediate frequency which is then filtered with a low pass filter  226  and then input to a digital logic block  228 . Similarly, the Q-channel is derived through the use of a down converter  230  which receives the input signal from node  220  and a clock signal from the clock recovery block  224  shifted in phase by 90° by a quadrature phase shift block  234 . This provides the Q-signal to a low pass filter  236 , the output of which is input to the digital logic block  228 . This provides a recovered data output therefrom. The digital logic block  228  provides the data processing operation wherein filtering is facilitated and data sampling is facilitated, as will be described in more detail hereinbelow. This where the large portion of the demodulation occurs at the baseband.  
         [0023]     Referring now to  FIG. 3 , there is illustrated a more detailed diagram of the receiver  104 . The output of the down converter  232  is input to an anti-aliasing filter  302  and, similarly, the output of the down converter  230  is input to the input of an anti-aliasing filter  304 . These operate in the analog domain. The output of the anti-aliasing filter provides analog IF which is input to an analog-to-digital converter  306  to provide a digital IF output on a digital bus  308 . The anti-aliasing filter  304  provides an analog IF for input to an analog-to-digital converter  310  to provide a digital IF output on a bus  312 . The bus  308  is input to an I-channel demodulator  314  and the digital bus  312  is input to a Q-channel demodulator  316 . The demodulators  314  and  316  each have the band-pass filter disclosed herein for providing the adjacent channel rejection while accounting for ISI interference such that the signal on the I-channel and the signal on the Q-channel can be recovered.  
         [0024]     Referring now to  FIG. 4 , there is illustrated the half-sine symbol for the O-QPSK signal for a single bit. It can be seen that a bit period of 1.0 microsecond is illustrated with a half-sine signal occupying a T c  of 0.5 microsecond. In time, the O-QPSK symbol equation is:  
                 s   ⁡     (   t   )       =     sin   ⁡     (       π   2     ·     t   Tc       )         ⁢     
     ⁢     where   ⁢     :               (   1   )               Tc   =     0.5   ⁢   μs             (   2   )             
 
 Thus, the half-sine signal will be a zero value at 0.0 μs, a maximum at 0.5 μs and a minimum at 1.0 μs. 
 
         [0025]     Referring now to  FIG. 5 , there is illustrated a plot of the power spectral density for the half-sine symbol of  FIG. 4 . This is illustrated by a solid line  502  that has a main lobe from 0.0 to 1.5 MHz, a secondary lobe  504  from 1.5 to 2.5 MHz and decreasing lobes each 1 MHz thereafter. In general, in order to reconstitute this signal, a filtering scheme should be utilized to recover substantially all of the energy in all of the lobes. The O-QPSK power spectral density is defined as follows:  
                    S   ⁡     (   f   )            =       1   π     ·       1     4   ·   Tc             (     1     4   ·   TC       )     2     -     f   2         ·     cos   ⁡     (     2   ⁢     π   ·   f   ·   TC       )                 (   3   )             
 
 Conventional demodulators will filter the signal  502  at some frequency between 1.5 to 2.5 MHz in order to recover substantially all of the transmitted power. Typically, some type of matched filter will be utilized. However, it can be seen that a carrier centered about a frequency  510  at 1.0 MHz from the center frequency of the transmitted symbol will be well within the main lobe, thus having the potential to contribute noise due to adjacent channel interference. This is a conventional adjacent channel of the type referred to a Blue Tooth. This utilizes Gaussian Frequency Shift Keying (GFSK) which basically involves passing the input signal through a Gaussian filter and then through a simple FSK subsystem. This will result in the following relationship:  
               g   ⁡     (   t   )       =       1         2   ⁢   π       ⁢   σ   ⁢           ⁢   T       ⁢     exp   ⁡     (       -     t   2         2   ⁢     σ   2     ⁢     T   2         )                 (   4   )               σ   =         ln   ⁡     (   2   )           2   ⁢   π   ⁢           ⁢   BT               (   5   )             
 
 If this signal is present centered 1.0 MHz from the signal of interest, it can result in a sufficient amount of energy being within the filter band if the filter band is between 1.5 to 2.5 MHz. In the present disclosed embodiment, his center frequency  510  has modulation associated therewith that will occupy a bandwidth from 0.5 MHz to 1.5 MHz. Thus, it would be desirable to filter the signals such that all of the energy from 0.0 to 0.5 MHz is recovered, which is substantially 80% of the energy, while rejecting energy above 0.5 MHz, such that substantially all the energy that would be associated with the adjacent channel would be rejected. However, as will be described hereinbelow, a consideration for this filtering is the ISI that might result within the recovered signal. This is a function of the filtering. In general, to reduce ISI, an ideal pulse shape would have zeros in the impulse response that would go through zero at equally spaced intervals that are multiples of the sampling interval. 
 
         [0026]     One type of filtering that can be utilized is that illustrated in  FIG. 6  and satisfies the criteria to provide zero crossings at substantially intervals of the sampling interval. In  FIG. 6 , the unfiltered power spectral density (PSD) response  502  is subjected to a raised-cosine filter, this is illustrated with a filter shape  602 . In general, this has a fairly flat in-band response with a slight roll-off at the corner frequency and then a very sharp roll-off to provide a pseudo brick-wall filter response at 0.5 MHz. A raised cosine filter is defined as a filter with a specific characteristic that no intersymbol interference at the sample times of adjacent signaling intervals. The raised cosine response will be as follows:  
                 H   ⁡     (   f   )       =   1     ,     f   &lt;     f   o               (   6   )                   H   ⁡     (   f   )       =     ·       1   +     cos   ⁡     (       π     2   ⁢           ⁢   α       ·     (       f     f   o       -   1   +   α     )       )         2         ,     f   ∈     [         f   o     ·     (     1   -   α     )       ;       f   o     ·     (     1   +   α     )         ]               (   7   )                   H   ⁡     (   f   )       =   0     ,     f   &gt;       f   o     ·     (     1   +   α     )                 (   8   )             
 
         [0027]     The resultant channel response in the frequency domain is illustrated in  FIG. 6   a . In  FIG. 6   a , it can be seen that the flat pass-band response for the filter will basically have a shape from 0.0 to 0.5 MHz that basically tracks the PSD response  502  from the original PSD response. Thus, the PSD of the channel response after filtering will not be flat in band. To further refine this, the embodiment of  FIG. 7  is referred to. In  FIG. 7 , the PSD response  502  is subjected to a filter that is a combination of a raised-cosine filter with an in band amplitude adjustment where it can be seen that the in band portion of the filter below 0.5 MHz has a gain associated therewith such that it will increase the response at 0.5 MHz and then roll off sharply at 0.5 MHz. The in-channel PSD response is illustrated in  FIG. 7   a , where it can be seen that the in band response is relatively flat.  
         [0028]     For the filter response of  FIG. 7  with the dotted line, the brick wall-like spectrum will be defined as follows. As a first approximation, the filter&#39;s frequency response is:  
                 H   ⁡     (   f   )       =     1     S   ⁡     (   f   )           ,     f   &lt;     f   o               (   9   )                   H   ⁡     (   f   )       =   0     ,     f   &gt;     f   o               (   10   )             
 
 This is basically the inverse of the PSD for the half-sine symbol. This will account for the roll off of the PSD energy over the low frequency flat portion of the raised cosine filter response. 
 
         [0029]     For practical implementation of the filter, to provide a channel response illustrated in  FIG. 7   a , the proposed transfer function, H(f), of Equations 9 and 10 can be combined with a raised cosine response:  
                 H   ⁡     (   f   )       =     1     S   ⁡     (   f   )           ,     f   &lt;       f   o     ·     (     1   -   α     )                 (   11   )                   H   ⁡     (   f   )       =       1     S   ⁡     (   f   )         ·       1   +     cos   ⁡     (       π     2   ⁢           ⁢   α       ·     (       f     f   o       -   1   +   α     )       )         2         ,     f   ∈     [         f   o     ·     (     1   -   α     )       ;       f   o     ·     (     1   +   α     )         ]               (   12   )                   H   ⁡     (   f   )       =   0     ,     f   &gt;       f   o     ·     (     1   +   α     )                 (   13   )             
 
 The resulting channel response in  FIG. 7   a  will therefore be that of a raised cosine response:  
                 H   ⁡     (   f   )       =   1     ,     f   &lt;       f   o     ·     (     1   -   α     )                 (   14   )                   H   ⁡     (   f   )       =       1   +     cos   ⁡     (       π     2   ⁢           ⁢   α       ·     (       f     f   o       -   1   +   α     )       )         2       ,     f   ∈     [         f   o     ·     (     1   -   α     )       ;       f   o     ·     (     1   +   α     )         ]               (   15   )                   H   ⁡     (   f   )       =   0     ,     f   &gt;       f   o     ·     (     1   +   α     )                 (   16   )             
 
 The results for a value of α=0.2 are illustrated in  FIG. 8 , wherein the unfiltered O-QPSK channel is illustrated with a PSD response  802 , the proposed base band filter response is illustrated with a filter response  804  and the filter O-QPSK channel is illustrated with a filter response  806 . 
 
         [0030]     The Fourier transformer of the filtered response will result in the channel time response therefor. For a single half symbol,  FIG. 9  illustrates the channel response for two cases, one for processing the signal through a standard raised cosine filter, which results in a response  902  and the other for processing through the filter response of  FIG. 7 . For the response  904 , it can be observed that the zeros for the timed response occur at exact intervals of 2 T c , allowing for minimum ISI with the response of  FIG. 7 , whereas the standard raised cosine filter results in only approximate zero-crossings. For example, at 1.0 μs, the response  904  will have a zero at a point  906 . However, the “null” of the response  902  will occur at a point  908  that is not quite at 1.0 μs. Each of the nulls in the response  902  will be slightly off from the exact 2.0 T c . Thus, as will be described hereinbelow with respect to the discussion of  FIG. 10 , each symbol is sampled at one microsecond time increments and, therefore, the response  904  will insure that the amount of signal contributed to a subsequent sample is substantially zero at the sampling time thereof. For example, the signal sample occurring at 1.0 μs after the signal illustrated in  FIG. 9  will have a substantially zero contribution from the response  904  at the point of sampling, but the response  902  will provide a contribution at a point  912  that is approximately 23 dB down. The point  906  will be approximately 100 dB down, a considerable difference. Thus, the ISI between two adjacent symbols at 1.0 μs out will be approximately −23 dB. It can therefore be seen that by utilizing the inverse of S(f) in combination with the raised cosine filter function at the corner frequency, the attenuation in the time domain at the sampling points of each symbol is increased over the raised cosine filter function in and of itself.  
         [0031]     Referring now to  FIG. 10 , there is illustrated a channel time response for subsequent symbols. The signal, when modulated, is illustrated as a plurality of half-sine symbols. For a data sequence “110010” there will be two positive half-sine symbols, two negative half-sine symbols, a positive half-sine symbols and then a negative half-sine symbols. The first half-sine symbols, half-sine symbols  1002 , will have associated therewith a first channel time response that will occur at time t 0 . This will be substantially the sampling time thereof, such that the recovered data will be sampled at time t 0 . As described hereinabove, a time response for the half-sine symbol  1002  will result in a zero at 1.0 μs from t 0 , a zero at 2.0 μs, etc. A second symbol,  1006 , will be generated which will have a channel time response defined by a second waveform at t 1 . The time t 1  will occur at 1.0 μs, which, if the data stream is sampled at t 1 , this will result in the sampling occurring at 1.0 μs from t 0  such that the contribution from the time response of symbol  1002  will be zero. This will continue wherein a third symbol  1008  will have a sampling point at t 2 , which will occur at a zero for the second symbol  1006  and a zero for the first symbol  1002 . This will continue for the fourth symbol, symbol  1010 , having a sampling point at t 3  which, again, occurs at a zero associated with the channel time responses of symbols  1002 ,  1006  and  1008 . A fifth channel time response is illustrated for the next and fifth symbol,  1012 , which will be sampled at time t 4 , which, again, will occur at the zeros for the channel time responses for symbols  1002 ,  1006 ,  1008  and  1010 . The actual signal output would be the sum of all of the channel time responses for all of the symbols, wherein it can be seen that by sampling at the periods t 0 , t 1 , t 2 , t 3 , t 4 , etc., each sample will occur at the zero for adjacent and previous symbols such that ISI is substantially eliminated.  
         [0032]     The filter function is realized with a digital filter. This can either be a finite impulse response filter or an infinite impulse response filter. The finite impulse response filter (FIR) is one that is utilized in the present disclosure. FIR filters have typically been referred to as moving average filters, transversal filters and non-recursive filters. These are conventional filters. The time response for one FIR implementation is depicted in  FIG. 11  for an alpha value of 0.4.  
         [0033]     Referring now to  FIG. 12 , there is illustrated a more detailed diagram of the receiver and the digital portion thereof. The analog IF channel is input to an ADC  1202  which is then processed through a digital down converter  1204 . The output of the digital down converter provides a digital IF on a bus  1206  which is then processed through a digital channel filter  1208  that provides the band-pass function, this being a FIR filter. This will typically have associated therewith filter coefficients in a storage area  1210 , these defining the operation of the filter. Once filtered, the output is then subjected to digital sub-sampling and clock recovery in a block  1212 . Additionally, the digital down converter  1204  utilizes some of the clock recovery for a complex digital multiplication, this being well known. The output will provide a recovered clock and data to a digital comparator  1214  to provide the recovered data. This is for a single channel which can then be combined with the data from the output channel.  
         [0034]     Although the preferred embodiment has been described in detail, it should be understood that various changes, substitutions and alterations can be made therein without departing from the spirit and scope of the invention as defined by the appended claims.