Abstract:
A servo channel digitally processes the data read from a magnetic media. The channel uses both edges of a system clock to detect peaks and generates position error systems by an area-based automatic gear control loop. By altering the sample delay, the channel digitally, up-samples at higher rates without requiring a higher system clock.

Description:
RELATED APPLICATIONS 
   This application is a continuation of application Ser. No. 10/114,578, filed on or about Apr. 2, 2002 now U.S. Pat. No. 6,606,358, which is a continuation of application Ser. No. 09/605,133, filed on or about Jun. 27, 2000, now U.S. Pat. No. 6,430,238, which is a continuation of application Ser. No. 09/049,830, filed on or about Mar. 27, 1998, now U.S. Pat. No. 6,125,154, the contents of each of are incorporated herein by reference. 

   FIELD OF THE INVENTION 
   This invention relates to servo pulse detection and position error signal demodulation for information storage/retrieval devices, such as magnetic disk drives. 
   BACKGROUND OF THE INVENTION 
   Servo detection and demodulation are commonly used in disk/tape drives in which information is stored on multiple tracks on a storage medium. In order to increase the storage density of these devices, the tracks are placed closer together, resulting in a tighter tolerance specification for positioning the read/write head over the surface of the medium. In a magnetic disk drive, servo data are usually written on the storage medium once during the manufacture of the drive. The servo patterns typically contain gray-coded track/sector identification (ID) information as well as positioning error information. When read by a magnetic pickup head, these data patterns present themselves as analog waveforms corrupted by electronics and media noise. The servo pulse detection circuit converts the analog pulses in the gray-code ID field of the servo pattern into clearly distinguishable digital pulses so that the information can be further processed using simple logic circuits. A servo error demodulator circuit determines the positioning error of the head relative to the center of the nearest track the head is located on. Conventional servo pulse detectors are typically designed using analog peak detectors similar to the conventional peak detector circuit used for the main data channel in magnetic disk. Integrating the servo channel and the main data channel on a monolithic silicon chip is relatively simple and has been a cost effective solution. However, with the advent of digital maximum likelihood channels which improves the recording density of magnetic disk drives, the main data channel circuitry becomes predominantly digital. Implementing the servo channel using digital circuitry thus become more desirable to ease the integration of the servo and main data channels. 
   It is desirable to provide a digital circuit technique for servo pulse detection and servo error demodulation which are compatible with the circuit techniques used in a digital read channels. 
   SUMMARY OF THE INVENTION 
   Several difficulties arise when performing digital pulse peak detection in digital domain. The main problem is that discrete time signal processing introduces a time quantization effect that can be reduced only through using a higher system sampling rate. The present invention uses both system clock edges to perform digital pulse peak detection to mitigate the inaccuracies caused by discrete time signal processing. 
   The present invention includes an area-based automatic gain control loop. This provides a very desirable feature of generating position error signals that are already normalized and independent of the incoming input frequency spectra when, for example, the head moves across multiple recording zones. 
   The present invention also includes a differentiating band-pass filter for servo burst filtering, independent of the equalizing filter used for servo pulse detection. The differentiation characteristic of this filter enables the accuracy of position error signal (PES) demodulation to be independent of the offsets in the analog front end circuits of the channel. The bandpass characteristic of the filter removes much of the noise in the PES bursts, resulting in a higher accuracy in PES demodulation in the presence of wide band input noise. 
   In the following description, we will use the term “digital data” or “digital vector” to imply a group of related digital signal bits (e.g., a digital bus, or a group of related digital buses) that represents an analog signal in digital domain. The term “digital signal” refers to a single bit digital line. 
   The input to the servo channel of the present invention is a first analog signal read back from the servo data field of a storage media. The servo data field contains a synchronization field, followed by a gray-code ID field and then a multiple of position error burst fields. 
   The servo channel also includes a programmable gain amplifier (PGA) to amplify the first analog signal to a second analog signal. An analog filter filters the second analog signal to provide a third analog signal. An analog to digital converter (ADC) digitizes the third analog signal to provide a first digital data. A digital differentiator receives a first digital data and provides, in response thereto, a second digital data. A digital up-sampler processes the second digital data to provide a plurality of digital data, each of which is an interpolated version of the second digital data at a different sampling delay. The digital up-sampler provides a higher equivalent sampling rate of the system without actually operating any circuits at a higher clock rate. An absolute value function circuit rectifies the interpolated digital data and sums them together to provide a fourth digital data. A digital area-based gain control unit (AGU) compares the signal level of the fourth digital data against a target value and generates a fifth digital data which controls the gain setting of the PGA. The AGU adjusts the gain of the PGA until the signal level of the fourth digital data achieves a certain target value. The signal path starting from the first analog signal to the fifth digital data forms an automatic gain control loop. This loop is active during the synchronization field of the servo loop. The gain of the PGA is frozen after the synchronization field. 
   A programmable coefficient digital FIR filter equalizes the first digital data to provide a sixth digital data. A digital peak detector processes the sixth digital data to provide a servo pulse signal and an optional pulse polarity signal. The FIR filter and the digital peak detector is used to provide a cleanly detected gray-code ID pulses for further servo ID detection by external control logic. 
   A digital area integrator integrates the fourth digital data to provide a plurality of digital outputs representing the servo position error signal (digital PES). This digital PES data can be read directly by an external servo DSP unit outside of this invention. An optional digital to analog converter (DAC) array converts the digital PES data back to analog PES signals to provide compatibility for back end servo processor systems that expect to receive the demodulated PES signals in analog form. 
   The digital area gain control unit comprises a first integrator which substantially integrates every half cycle of the servo sync field section of the fourth digital data. This is achieved by making the half cycle period in the servo sync-field substantially equal to an integer multiple of the sampling clock period. The half cycle integrated value is compared against a target level and a difference value is generated and referred to as the gain error data. The gain error data is further accumulated by a second integrator to produce the gain control data for the PGA. The second integrator includes a saturator to prevent overflow or underflow of the gain control data. 
   The digital peak detector comprises a differentiator, a threshold detector and a zero-crossing detector. The differentiator converts peaks in the incoming data into zero-crossings in its outgoing data. The threshold detector produces a valid-positive-peak data indicator any time the incoming signal is greater than a certain positive threshold, and a valid negative peak indicator when the signal is below a certain negative threshold. The zero-crossing detector produces a negative-servo-pulse output and a positive-servo-pulse output. The negative servo-pulse is asserted when a positive transitioned zero crossing is detected and the valid-negative-pulse output is asserted. The positive servo-pulse is asserted when the negative transitioned zero crossing is detected and the valid-positive-pulse output is asserted. An optional OR gate combines the positive servo-pulse and the negative servo-pulse signals together to provide a composite servo-pulse output. An optional set-reset flip-flop has its set and reset inputs controlled by the negative-servo-pulse and the positive-servo-pulse to provide an output indicating the original polarity of the servo pulse for the composite servo-pulse output. A multiplexer selects either the separated negative/positive servo pulse signals, or the composite and polarity signals as the output of the servo pulse detector. 
   The digital area integrator for PES demodulation integrates the PES burst field section of the fourth digital data each time the burst gate control signal is asserted. The integration length is the smaller of the burst gate assertion time period and a programmed burst count value. The integrated value is sequentially loaded into a plurality of registers upon every deassertion of the burst gate signal. Under normal operation, the burst gate assertion time period in number of the servo system clock preferably is longer than the programmed burst count value. The user may also program the burst count value so that the total integration time substantially covers an integral multiple of the servo PES burst cycles for improved PES demodulation accuracy. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a block diagram illustrating a digital servo channel in accordance with the present invention. 
       FIG. 2  is a graphical view illustrating a typical servo read back waveform from a conventional magnetic disk drive. 
       FIG. 3  is a block diagram illustrating a digital area-based gain control circuit of FIG.  1 . 
       FIG. 3   a  is a block diagram illustrating an exemplary gain-error generator of FIG.  3 . 
       FIG. 3   b  is a block diagram illustrating an exemplary gain-integrator of FIG.  3 . 
       FIG. 4  is a graph illustrating the desired magnitude transfer function for the differentiating band-pass filter of FIG.  1 . 
       FIG. 5  illustrates an implementation of the interpolator array of  FIG. 1  for m=2, which is the preferred embodiment of the present invention. 
       FIG. 6  is a block diagram of the absolute value summer circuit of FIG.  1 . 
       FIG. 7  is the block diagram illustrating the digital peak detector of  FIG. 1  in accordance with the present invention. 
       FIG. 8  is a timing diagram illustrating the internal timing of the peak detector circuit of FIG.  7 . 
       FIG. 9  is a block diagram illustrating the digital area integrator of  FIG. 1  in accordance with the present invention. 
       FIG. 9   a  is an illustration of the internal timing of the digital area integrator of FIG.  9 . 
   

   DETAILED DESCRIPTION 
   Referring to  FIG. 1 , there is shown a schematic block diagram illustrating a digital servo channel  100  in accordance with the present invention. The digital servo channel  100  includes an automatic gain controller  101 , a digital position error demodulator  103 , and a servo gray code pulse detector  105 . 
   The automatic gain controller  101  includes a programmable gain amplifier (PGA)  102 , an analog filter  104 , an analog-to-digital converter (ADC)  106 , a digital differentiator  108 , an interpolator array  114 , an absolute value summer circuit  116 , and a digital gain control circuit  118 . An analog input signal  131  from a transducer (not shown) is applied to an input of the programmable gain amplifier (PGA)  102 , which provides an amplified analog signal  133  to the input of the analog filter  104  in response to the input signal  131  and to a gain control vector  145 . The analog filter  104  provides a filtered analog signal  135  to the input of an analog-to-digital converter (ADC)  106 . In some channels  100 , the analog filter  104  need not be present, and, in these channels, the analog signals  133  and  135  are identical. The ADC  106  digitizes the filtered analog signal  135  to provide a raw digital vector  137 , which is a digital representation of the analog signal  135 , to the digital differentiator  108 . The digital differentiator  108  preferably has the transfer function shown in FIG.  4 . In response to the raw digital vector  137 , the digital differentiator provides a digital differentiated filtered vector  139  to the interpolator array  114 . The interpolator array  114  processes the digital vector  139  to provide a plurality of interpolated digital vectors  141 - 1  through  141 - m . The digital vectors  141 - 1  through  141 - m  form a close representation of the digital vector  139  at different sampling times. Each interpolated digital vector  141  is indicative of an upsampled value representative of the differentiated filter signal  139  at a different sample time. One of the interpolated digital vectors  141 - 1  through  141 - m  may be a delayed version of the digital vector  139  to simplify the hardware requirements of the interpolator array  114 . 
   The absolute value summer circuit  116  rectifies the interpolated digital vectors  141 - 1  through  141 - m  and arithmetically sums them together to provide a digital rectified vector  143 , which is indicative of the rectification and summing of the absolute values of the interpolated digital vectors  141 - 1  through  141 - m . The digital rectified vector  143  may be, for example, indicative of the half cycle area of the sync field burst when the channel  100  is processing the servo sync field region. In response to the digital rectified vector  143 , the digital gain control circuit  118  provides the digital gain control vector  145  to control the gain setting of the PGA  102 . When the servo channel  100  is in a gain acquisition mode, the digital gain control vector  145  is adjusted until the magnitude of the digital rectified vector  143  reaches a predetermined level. 
   The servo gray code pulse detector  105  includes a programmable coefficient digital finite impulse response (FIR) filter  110  and a digital peak detector  112 . The programmable coefficient digital FIR filter  110  equalizes the raw digital vector  137  to provide a digital vector  151 . The digital peak detector  112  processes the digital vector  151  to provide digital signals  153  and  155 . The digital peak detector  112  can be configured so that the digital signals  153  and  155  either include the servo-pulse signal and the servo pulse-polarity signal, respectively, or include the positive servo-pulse signal and the negative servo-pulse signal, respectively. 
   The digital position error demodulator  103  includes a digital area integrator  120  and a digital-to-analog converter (DAC) array  122 . The digital area integrator  120  integrates the digital rectified vector  143  to generate a plurality of digital vectors  147 - 1  through  147 - n  which represent the servo position error signals (digital PES vectors). The digital to analog converter (DAC) array  122  converts the digital PES vectors into analog PES signals  149 - 1  through  149 - n  to provide backward compatibility for back-end systems that receive the demodulated PES signals in analog form. Of course if the back-end system receives digital PES signals, the digital position error demodulator  103  need not include the DAC array  122 . 
   Referring to  FIG. 2 , there is shown a graphical view illustrating an example of a servo read back waveform of a conventional servo patterns used in magnetic disk drives. The servo read-back waveform typically includes a single frequency servo sync-field  201 , a gray-coded servo track/sector ID field  203 , an address mark or gap field  202  separating the sync-field  201  from the ID field  203 , and also includes a plurality of single frequency servo position error burst signals  204 - 1  through  204 - n . The servo sync field  201  provides a training field for a servo channel (not shown) to adjust its gain control loop. During the track/sector ID field, the servo channel converts received analog pulse patterns into an unambiguous train of digital pulse patterns with low error rate for further processing by back-end controller circuits which only handle digital pulses. The gap field or address mark indicates to the back-end controller the start of the ID field. Typically, the digital pulses to be generated during ID field processing are designed to be located at the peaking instances of the input analog waveform. Thus, a peak detector is commonly used to perform servo pulse detection. Other servo schemes may use the zero crossings of the analog waveform to encode the digital pulse position. In this case, servo pulse detection can still be performed using a peak detector by first differentiating the incoming signal to convert zero-crossings into analog pulse peaks. 
   The head tracking information is derived from the servo PES fields. Typically, several servo burst fields are written on the disk in a staggered fashion so that the read back amplitude of each one of them will be different and depends on the positioning of the head. A common scheme used in the art is to use four servo PES bursts commonly referred to as the A, B, C and D bursts. By reading the magnitude of burst A, B, C and D, a back-end servo processor can make the correction to guide the head on track. The servo burst demodulator converts the single tone sinusoidal like burst signals from the read back waveform into clean DC signals for representing the magnitude of burst A, B, C and D respectively. 
   Referring to  FIG. 3 , there is shown a schematic block diagram illustrating the digital area-based gain control circuit  118 . The digital gain control circuit  118  includes an N-cycle integrator  302 , a gain-error generator  304 , and a gain integrator  306 . The N-cycle integrator  302  substantially integrates a half cycle of the digital rectified signal  143  to generate an area signal  331 . This can be easily achieved by making the half cycle period substantially equal to an integer multiple of the sampling clock period. The half cycle integrated value represents the area of a half cycle in the servo sync-field. The gain-error generator  304  compares the half cycle area value to a predetermined value and generates a gain error signal  333 , which is accumulated by the gain integrator  306  during sync-field acquisition to produce the digital gain control vector  145 . 
   Referring to  FIG. 3   a , there is shown a block diagram illustrating an exemplary gain-error generator  304 , which includes a saturator  308 , a subtractor  310 , and a multiplier  312 . The subtractor  310  subtracts the half cycle area signal  331  from a pre-selected target value  339  to produce a raw gain error signal  335 . The saturator  308  is a minimum-maximum limiter which processes the raw gain error signal  335  to generate a modified gain error signal  337  having a value that is within a range less than the range of the values of the raw gain error signal  335 . The multiplier  312  multiplies the modified gain error signal  337  with a gain-error scaling value  341  to generate the final gain error signal  333 . The gain-error scaling value  341  is programmed or hardwired to achieve the desired gain acquisition tracking bandwidth. Pipeline delays may be added to the gain-error generator  304  to increase the speed of the gain-error generator  304 . 
   Referring to  FIG. 3   b , there is shown a block diagram illustrating an exemplary gain integrator  306  of FIG.  3 . The gain integrator  306  includes an adder  314 , a saturator  316 , and a register  318 . During normal accumulation, the adder  314  adds the gain error signal  333  to the digital gain control signal  145  to produce a next gain control signal  351 . The saturator  316  limits the range of the next gain control signal  351  to generate a range-limited gain control signal  353 . The register  318  receives the range-limited gain control signal  353  and transfers out the digital gain control signal  145  in the next accumulator update cycle. The saturator  316  prevents overflow and underflow of the arithmetic operation involved in integration. 
   Referring to  FIG. 4 , there is shown a graph illustrating the magnitude transfer function for the differentiating band-pass filter (DIFF)  108 . Since the sync-field as well as the PES burst fields are single tone frequency pattern, it is advantageous to use a bandpass filter to pass the desired burst signals and reject other noise components as much as possible. Towards this end, a differentiating bandpass filter may be used because of its extra capability of rejecting DC offset value from the input signal. The desired magnitude transfer function is shown in  FIG. 4  The transfer function is zero at DC, peaks at around the frequency of the sync/PES field fundamental frequency, and drops to a low value after twice the peaking frequency. The filter simultaneously rejects the DC component as well as the high frequency noise component in the digitized signal  139 . The filter may be, for example, an FIR filter. For a simplified implementation, the filter may be an FIR filter with fixed binary coefficients of simple powers of two. For a servo channel operating at a sampling clock of approximately 8 times the burst frequency, an FIR filter with coefficients having relative values of 1,2,1,0,−1,−2,−1 may be used. 
   Referring to  FIG. 5 , there is shown a block diagram illustrating an exemplary interpolator array  114 . For clarity, an interpolator array  114  with m=2 is shown. The interpolator array  114  includes a delay/buffer  402 , a delay circuit  404 , and an adder  406 . General signal interpolation can be performed using FIR filters of appropriate coefficients and is well known in the art. For a simple hardware implementation, the delay/buffer  402  provides the first interpolated value  141 - 1  which equals a delayed/buffered value of the digital vector  139 . The delay circuit  404  provides a delayed digital vector  139  to the adder  406 , which adds the delayed vector to the digital vector  139  to generate the second interpolated value  141 - 2 . The second interpolated value  141 - 2  is an equally weighted average of consecutive sample points. The average may be generated by an FIR filter with filter coefficients of (0.5, 0.5), which is a linear interpolation scheme. A higher level of interpolation with m&gt;2 is achieved with linear interpolation with general coefficients of (c, 1−c). Higher order interpolation with more FIR coefficients may be used to improve the interpolation result. 
   Referring to  FIG. 6 , there is shown a block diagram illustrating the absolute value summer circuit  116 , which includes absolute value generators  802 - 1  through  802 - m , and a summer  804 . The absolute value generators  802 - 1  through  802 - m  generate respective absolute value signals  831 - 1  through  831 - m , which are the absolute value of respective interpolated digital vectors  141 - 1  through  141 - m  provided by the interpolator array  114 . The summer  804  sums the absolute value signals  831 - 1  through  831 - m  together to produce the digital rectified vector  143 . A number m of interpolated digital vectors  141  greater than 1 reduces the variation in the absolute-area integration values due to uncertain phase relationship between the incoming analog input signal  131  and a digital system clock (not shown). 
   Referring to  FIG. 7 , there is shown a block diagram illustrating the digital peak detector  112 . Referring to  FIG. 8 , there is shown a timing diagram illustrating the timing of the digital peak detector  112 . The digital peak detector  112  includes a late peak detector  702 , a threshold detector  704 , a zero-crossing detector  706 , a differentiator  708 , delay circuits  710 ,  712 ,  714 ,  716 , and  718 , an invertor  720 , AND gates  722 ,  724 ,  726 ,  728 ,  730 , and  732 , and OR gates  734  and  736 . The digital peak detector  112  performs signal peak detection in a digital domain as opposed to an analog domain. The output signals of the digital peak detector  112  are pulses similar to those of an analog peak detector. Because the digital peak detector  112  operates at a finite clock operating frequency, the digital output pulses occur on the sampling clock edges. This introduces time quantization effects, reducing the accuracy of recovered peak position compared to an analog peak detector. To mitigate the time quantization effect, the peak detector circuit  112  uses both the rising and falling edges (i.e., both clock phases) of the system sampling clock to generate the output pulses. This effectively doubles the sampling rate of the system to improve the precision in the recovery of the peak positions in the incoming signal. 
   The threshold detector  704  produces QPP and QNP signals. The QPP signal is asserted any time that the input digital vector  151  exceeds a programmed positive threshold PTHR. Similarly, the QNP signal is asserted any time the input digital vector  151  is below a programmed negative threshold NTHR. The QNP and QPP signals are used to qualify only peaks that exceeds the specified threshold NTHR and PTHR, respectively, to reject unwanted peaks around the base-line of the input digital vector  151 . A peak in the input digital vector  151  is typically detected by detecting a zero crossing in the input digital vector  151 . This is typically done by first differentiating the input digital vector  151  so that peak locations become zero-crossing locations. The zero-crossing detector  706  detects zero-crossing for both positively going and negatively going signal transitions. The state equations of the digital peak detector  112  are as follows: 
   The state equations for the threshold detector  704  are:
 
 QNP[n]=X[n]&lt;*NTHR   (1) 
 
 QPP[n]=X[n]&gt;*PTHR   (2) 
 
   The state equations for the differentiator  708  are:
 
 Z[n]=X[n]−X[n− 1]  (3) 
 
   The state equations for the zero-crossing detector  706  are either:
         equations (4a) and (5a)
 
 PX[n]= ( Z[n]&gt;= 0)  AND  ( Z[n− 1]&lt;0)  (4a) 
 
 NX[n]= ( Z[n]&lt;= 0)  AND  ( Z[n− 1]&gt;0)  (5a) 
   or equations (4b) and (5b)
 
 PX[n]= ( Z[n]&gt; 0)  AND  ( Z[n− 1]&lt;=0)  (4b) 
 
 NX[n]= ( Z[n]&lt; 0)  AND  ( Z[n− 1]&gt;=0)  (5b) 
       

   The state equations for the valid/qualified zero-crossing determined by the AND gates  722  and  724  are:
 
 QNX[n]=PX[n] AND QNP[n− 1]  (6) 
 
 QPX[n]=NX[n] AND QPP[n− 1]  (7) 
         where in equations (1) through (7),   1. the operator, “&lt;*” can be either less-than “&lt;” or less-than-or-equal-to “&lt;=”;   2. the operator “&gt;*” can be either greater-than “&gt;” or greater-than-or-equal-to “&gt;=”;   3. X[n] is the incoming input vector  151  from the FIR filter  110 ;   4. Z[n] is a difference input vector,   5. PX[n] indicates the occurrences of all negative peaks of X[n] or all positive going zero-crossings of Z[n];   6. NX[n] indicates the occurrences of all positive peaks of X[n] or all negative going zero-crossings of Z[n];   7. QNX[n] indicates the presence of a positive peak in X[n] that exceeds the specified positive threshold PTHR;   8. QPX[n] indicates the presence of a peak in X[n] that exceeds the specified negative threshold NTHR.       

   The state equations (1) through (7) provide a simple means of implementing the digital peak detector  112 . The digital peak detector  112  operates on the system sampling clock. Hence, the QNX signal changes value only after the triggering clock edge. To reduce the time quantization effect, the other clock phase of the system clock is also utilized. To do this, the pulse peak position is further determined to occur either early in the clock cycles or late in the clock cycles. The following state equations determine the position of the pulse peak: 
   The state equations for the early/late peak location detector  702  and the inverter  720  are:
 
 Late[n]= ( X[n+ 1 ]&gt;*X[n− 1])  AND  ( X[n]&gt;* 0) 
 
 OR  ( X[n+ 1 ]&lt;*X[n− 1])  AND  ( X[n]&lt;* 0)  (8) 
 
 Early[n]=NOT Late[n]   (9) 
 
   The state equations for the pulse shifting of the AND gates  726 ,  728 ,  730 , and  732 , and the OR gates  734  and  736  are:
 
 NX   —   E[n+ 0.5 ]=Early[n− 1 ] AND QNX[n]   (10) 
 
 PX   —   E[n+ 0.5 ]=Early[n− 1 ] AND QPX[n]   (11) 
 
 NX   —   L[n+ 1 ]=Late[n− 1 ] AND QNX[n]   (12) 
 
 PX   —   L[n+ 1 ]=Late[n− 1 ] AND QPX[n]   (13) 
 
 NX=NX   —   E[n+ 0.5 ] OR NX   —   L[n+ 1]  (14) 
 
 PX=PX   —   E[n+ 0.5 ] OR PX   —   L[n+ 1]  (15) 
         where in equations (8) through (15),   1. the 0.5 in NX_E[n+0.5] and PX_E[n+0.5] indicates that both signals are latched on the second phase of the system clock.   2. the 1 in NX_L[n+1] and PX_[n+1] indicates that both signals are latched on the main (first) phase of the system clock.   3. NX is the final positive pulse peak output of the peak detector  112 .   4. PX is the final negative pulse peak output of the peak detector  112 .       

   The state equations (8) through (15) shift the output pulses by a half clock period relative to the sample point depending on whether the actual signal peak would have occurred early or late relative to the digital peak sample point, as illustrated in FIG.  8 . In this case, if the actual peak would have occurred after the digital peak sample point X 2  of  FIG. 8 , the output pulse lines up with the system clock and is sent out on the next system clock cycle. If the actual peak position would have occurred before the digital sample peak position, the output pulse is latched earlier by the second phase of the system clock. 
   To obtain the more common servo output format of a composite pulse output (occurrence of either positive or negative peaks) and peak polarity output, the digital peak detector  112  may include a simple circuit (not shown) comprising an OR gate and an RS flip-flop. The additional OR gate provides the composite pulse output as the OR of the NX and PX signals. The output of the RS flip-flop provides the pulse polarity output. The Reset and Set inputs of the RS flip-flop are separately connected to the NX and PX signals. 
   Referring to  FIG. 9 , there is shown a schematic block diagram illustrating the digital area integrator  120 , which includes a burst integrator  902 , a sequencer  910 , and a plurality of PES holding registers  912 - 1  through  912 - n . The digital area integrator  120  demodulates the digital rectified vector  143  to generate the servo position error vectors  147 . The sequencer  910  generates a reset signal  914  and a plurality of load signals  916 - 1  through  916 - n  in response to a servo gate (BCNT) signal  918  and a burst gate (BGATE) signal  920 . 
   The burst integrator  902  includes an adder  904 , an AND gate  906 , and an accumulator register  908 . The burst integrator  902  integrates the incoming rectified signal  143  when the reset signal  914  is deasserted, and resets the accumulation register  908  when the reset signal  914  is asserted. At the end of every integration sequence, the sequencer  910  simultaneously asserts one of the plurality of load signals  916 - 1  through  916 - n , which enables loading of the value at the output of the accumulator register  908  before it is reset. The PES holding registers  912 - 1  through  912 - n  are sequentially loaded with the demodulated PES values of the corresponding servo burst field. The output of the registers  912 - 1  through  912 - n  provide the respective digital vectors  147 - 1  through  147 - n , which may be read directly by a servo digital signal processing controller (not shown) or they can be converted in analog signals using the digital to analog converters  122  for backward compatibility to older servo systems that receive the demodulated signals in analog form. 
   Referring to  FIG. 9   a , there is shown a timing diagram illustrating the timing of integrate/load cycle of the PES signals by sequencer  910 . The sequencer  910  is enabled when the servo gate signal  918  is asserted. The sequencer  910  generates a synchronized integrate/reset sequence on the reset line  914  in response to the burst gate signal  920 . The integrate cycle lasts for a programmed number of system clock cycles. At the end of the first integration cycle for the first PES burst field, the integrate cycle is terminated, and the load signal  916 - 1  is asserted to allow loading of the integrated value of the burst integrator  902  into the first PES holding register  912 - 1 . Subsequent burst gate assertion/deassertion cycles enable more integration cycles, but the sequencer  910  directly loads the integrated values into other PES holding registers  912 - 2  through  912 - n  by sequential asserting the load signals  916 - 2  through  916 - n.