Abstract:
An oscillator formed from low cost discrete semiconductors and passive devices creates a linear periodic ramp of constant frequency with ramp slope based on an external voltage signal. Parameters are stable over a wide range of temperatures and variations of transistor parameters that normally degrade in extreme environments. The oscillator period can be phase and frequency synchronized to an external clock source over a wide range of frequencies. The oscillator ramp generator phase can be synchronized on a cycle by cycle basis for incorporation in power converters employing spread spectral EMI reduction techniques, multi-converter systems employing clock interleaving for distribution bus filter optimization, and resonant mode converters employing zero voltage switching techniques. Oscillator ramp rate is independent of frequency and can be synchronized to DC (inhibit) for use in ultra low power burst mode power conversion.

Description:
BACKGROUND 
     1. Technical Field 
     This disclosure is generally related to oscillators, which may, for example, be employed in power converters and other devices. 
     2. Description of the Related Art 
     Power converters are used to transform electrical energy, for example converting between alternating current (AC) and direct current (DC), adjusting (e.g., stepping up, stepping down) voltage levels and/or frequency. 
     Power converters take a large variety of forms. One of the most common forms is the switched-mode power converter or supply. Switched-mode power converters employ a switching regulator to efficiently convert voltage or current characteristics of electrical power. Switched-mode power converters typically employ a storage component (e.g., inductor, transformer, capacitor) and a switch that quickly switches between full ON and full OFF states, minimizing losses. Voltage regulation may be achieved by varying the ratio of ON to OFF time or duty cycle. Various topologies for switched-mode power converters are well known in the art including non-isolated and isolated topologies, for example boost converters, buck converters, synchronous buck converters, buck-boost converters, and fly-back converters. 
     In the interest of efficiency, digital logic technology is employing ever lower voltage logic levels. This requires power converters to deliver the lower voltages at higher currents level. To meet this requirement, power converters are employing more energy efficient designs. Power converters are also increasingly being located in close proximity to the load in as point of load (POL) converters in a POL scheme. These power converters must generate very low voltage or potential levels (e.g., less than 1V) at increasingly higher current levels (e.g., greater than 10 A). These relatively high current levels may be difficult to achieve with a single power converter. 
     Manufacturers are also increasingly employing POL schemes in light of the widely varying voltage requirements in modern systems (e.g., computer systems). A POL scheme may be easier to design and/or fabricate, take up less area, and/or produce less interference than employing multiple different power buses. The POL schemes typically employ one or two power buses with a number of POL regulators located close to specific components or subsystems to be powered, for example microprocessors, field programmable gate arrays (FPGAs), application specific integrated circuits (ASICs), volatile memory. The POL regulators adjust voltage or potential to supply localized busses feeding the specific components or subsystems. 
     Switched mode power converters typically include an oscillator or oscillator circuit which generates, provides or supplies a periodic, oscillating ramp voltage waveform to a gate timing and drive control for use in generating pulse width modulated control signals for controlling one or more switches of the converter circuit. It is desirable that the oscillator provide a linear ramp signal with ramp voltage rate of change (dv/dt) rate proportional to the supply input voltage while maintaining a fixed frequency over a wide range of frequencies and over a wide range of temperatures. Additionally, the oscillator may need to be synchronized with other oscillating waveforms and other frequencies, for example those from a system clock or another power converter. Further, it is typically desirable that the oscillator is relatively inexpensive to manufacture. 
     New approaches to related to oscillation circuits are desirable. 
     BRIEF SUMMARY 
     The approaches described herein to generate a periodic ramp oscillator signal may advantageously provide one or more of: 1) the oscillator is self oscillating and requires no start up input; 2) the slope of the ramp is modulated by an external signal and is linear over a wide range of operating conditions of input signal range &amp; temperature; 3) the oscillator signal output can be synchronized to an external clock signal input; 4) the oscillator signal output synchronizes to the external signal input up to two times the self oscillation frequency of the oscillator; and/or 5) method uses less complex, lower cost discrete semiconductor components, chip resistors and capacitors. 
     In contrast, existing approaches utilize more costly and complex integrated circuit devices, such as comparators/operational amplifiers with additional components or a custom designed complex integrated circuit. Existing approaches appear to lack all the features described above. Existing approaches also appear to fail to operate over an ultra wide synchronous frequency range and temperature range as desired, with external slope modulation of the ramp oscillator signal. Existing approaches use terrestrial technology. 
     An oscillator ramp generator may be summarized as including a voltage input terminal to receive an input voltage signal; a synchronizing signal input terminal to receive a synchronizing signal; a ramp timing resistance; a ramp timing capacitor which is charged by the input voltage signal through the ramp timing resistance; an input voltage resistor divider network coupled to the voltage input terminal and which produces a discharge trigger reference voltage that is proportional to a voltage of the input voltage signal; a comparator that compares a voltage across the ramp timing capacitor to the discharge trigger reference voltage; and a logic circuit that in response to a first state of a synchronization signal, discharges the ramp timing capacitor and that in response to a second state of the synchronization signal when the value of the voltage across the ramp timing capacitor is equal to the discharge trigger reference voltage, delays a discharging of the ramp timing capacitor. 
     The input voltage resistor divider network may include a first divider resistor and a second divider resistor, the input voltage resistor network coupled to provide the discharge trigger reference voltage the comparator. The input voltage resistor divider network may further include a temperature compensation resistor to compensate the discharge trigger reference voltage for temperature variation. The comparator may include a first comparator transistor and a second comparator transistor coupled as a differential pair of transistors, with a base of the first comparator transistor coupled as an inverting input to the input voltage resistor divider network to receive the discharge trigger reference voltage and a base of the second comparator transistor coupled as a non-inverting input to the ramp timing capacitor to receive the voltage across the ramp timing capacitor. The comparator may include a first comparator transistor and a second comparator transistor coupled as a differential pair of transistors. 
     The oscillator may further include a control transistor; a discharge controlling capacitor; a discharge termination transistor and a discharge controlling transistor, the control transistor responsive to at least an output of the comparator to selectively charge the discharge controlling capacitor from the voltage input terminal, the discharge controlling capacitor coupled to selectively cause the discharge controlling transistor and the discharge termination transistor to discharge the ramp timing capacitor. The discharge controlling transistor may selectively provide a path to a ground to at least approximately fully discharge the ramp timing capacitor to at least close to a zero voltage. The logic circuit may include: a synchronization capacitor coupled to the synchronization input terminal to receive the synchronization signal, a first diode coupled to cause the control transistor to be in a first state (e.g., ON) in response to a first state (e.g., rising edge of a pulse) of the synchronization signal and a second diode coupled to the control transistor cause the control transistor to be in a second state (e.g., OFF) in response to a second state (e.g., low level pulse) of the synchronization signal. 
     The oscillator may further include a level shifting diode electrically coupling a control transistor resistor to a ground to level shift the control transistor resistor reducing a voltage required to turn on the control transistor. The ramp timing resistance may include a first ramp resistor and a second ramp resistor electrically coupled in series with the first ramp resistor between the voltage input terminal and the ramp timing capacitor. 
     The oscillator may further include a buffer transistor; and a buffer resistor, the buffer transistor and buffer resistor coupled to form an emitter follower buffer amplifier between a voltage supply and a ground to provide a low source impedance driver. At least the comparator and the logic circuit may each be formed of a number of discrete semiconductor components. 
     A method of operating an oscillator may be summarized as including receiving an input voltage signal at a voltage input terminal; receiving a synchronizing signal at a synchronizing signal input terminal; charging a ramp timing capacitor via the input voltage signal through a ramp timing resistance; producing a discharge trigger reference voltage that is proportional to a voltage of the input voltage signal; in response to a value of a voltage across the ramp timing capacitor being equal to the discharge trigger reference voltage, discharging the ramp timing capacitor if a state of the synchronization signal indicates synchronization and delaying discharging of the ramp timing capacitor if the state of the synchronization signal does not indicate synchronization; and in response to the state of the synchronization signal indicating synchronization while the voltage across the ramp timing capacitor is less than the discharge trigger reference voltage, discharging the ramp timing capacitor. 
     Comparing the voltage across the ramp timing capacitor to the discharge trigger reference voltage may include supplying the discharge trigger reference voltage to a first transistor of a differential pair of transistors and supplying the voltage across the ramp timing capacitor to a second transistor of the differential pair of transistors. 
     The method may further include determining whether to trigger discharging of the ramp timing capacitor when the value of a voltage across the ramp timing capacitor is equal to the discharge trigger reference voltage. Determining whether to trigger discharging of the ramp timing capacitor when the value of a voltage across the ramp timing capacitor is equal to the discharge trigger reference voltage may include supplying the synchronization signal to a logic network comprising a synchronization capacitor, a first synchronization diode and a second synchronization diode, triggering the discharging of the ramp timing capacitor via the first synchronization diode in response to a first synchronization state and suppressing triggering of the discharging of the ramp timing capacitor via the second synchronization diode in response to a second synchronization state. 
     The method of may further include buffering an output voltage signal via a buffer amplifier. Buffering an output voltage signal via a buffer amplifier may include buffering the output voltage signal via an emitter follower buffer amplifier having a discrete transistor and resistor. Producing a discharge trigger reference voltage that is proportional to a voltage of the input voltage signal may include dividing the input voltage signal via an input voltage resistor divider network and temperature compensating the divided input voltage signal. 
    
    
     
       BRIEF DESCRIPTION OF THE SEVERAL VIEWS OF THE DRAWINGS 
       In the drawings, identical reference numbers identify similar elements or acts. The sizes and relative positions of elements in the drawings are not necessarily drawn to scale. For example, the shapes of various elements and angles are not drawn to scale, and some of these elements are arbitrarily enlarged and positioned to improve drawing legibility. Further, the particular shapes of the elements as drawn, are not intended to convey any information regarding the actual shape of the particular elements, and have been solely selected for ease of recognition in the drawings. 
         FIG. 1  is a schematic diagram of a power converter including a oscillator ramp circuit, according to one illustrated embodiment. 
         FIG. 2  is a functional block diagram of the oscillator circuit of  FIG. 1 , according to one illustrated embodiment. 
         FIG. 3  is a detailed electrical schematic diagram of the oscillator circuit of  FIGS. 1 and 2 , according to one illustrated embodiment. 
         FIG. 4  is a flow diagram of a method of operating the oscillator circuit of  FIGS. 1-3 , according to one illustrated embodiment. 
         FIG. 5  is a flow diagram of a method of operating the oscillator circuit of  FIGS. 1-3 , according to one illustrated embodiment, which may be implemented as part of performing the method of  FIG. 4 . 
         FIG. 6  is a flow diagram of a method of operating the oscillator circuit of  FIGS. 1-3 , according to one illustrated embodiment, which may be implemented as part of performing the method of  FIG. 4 . 
         FIG. 7  is a flow diagram of a method of operating the oscillator circuit of  FIGS. 1-3 , according to one illustrated embodiment, which may be implemented as part of performing the method of  FIG. 4 . 
     
    
    
     DETAILED DESCRIPTION 
     In the following description, certain specific details are set forth in order to provide a thorough understanding of various disclosed embodiments. However, one skilled in the relevant art will recognize that embodiments may be practiced without one or more of these specific details, or with other methods, components, materials, etc. In other instances, well-known structures associated with power conversion topologies have not been shown or described in detail to avoid unnecessarily obscuring descriptions of the embodiments. 
     Unless the context requires otherwise, throughout the specification and claims which follow, the word “comprise” and variations thereof, such as, “comprises” and “comprising” are to be construed in an open, inclusive sense, that is as “including, but not limited to.” 
     Reference throughout this specification to “one embodiment” or “an embodiment” means that a particular feature, structure or characteristic described in connection with the embodiment is included in at least one embodiment. Thus, the appearances of the phrases “in one embodiment” or “in an embodiment” in various places throughout this specification are not necessarily all referring to the same embodiment. Furthermore, the particular features, structures, or characteristics may be combined in any suitable manner in one or more embodiments. 
     As used in this specification and the appended claims, the singular forms “a,” “an,” and “the” include plural referents unless the content clearly dictates otherwise. It should also be noted that the term “or” is generally employed in its sense including “and/or” unless the content clearly dictates otherwise. 
     As used in the specification and the appended claims, references are made to a “node” or “nodes.” It is understood that a node may be a pad, a pin, a junction, a connector, a wire, or any other point recognizable by one of ordinary skill in the art as being suitable for making an electrical connection within an integrated circuit, on a circuit board, in a chassis or the like. 
     The headings and Abstract of the Disclosure provided herein are for convenience only and do not interpret the scope or meaning of the embodiments. 
       FIG. 1  shows a power converter  100 , according to one illustrated embodiment. The description of  FIG. 1  provides an overview of the structure and operation of the power converter  100 , which structure and operation are described in further detail with reference to  FIGS. 2-7 . 
     The power converter  100  may, for example, take the form of a DC/DC power converter to convert (e.g., raise, lower) DC voltages. The power converter  100  may, for example, include an output inductor L out  electrically coupled to an output terminal +VOUT, a first active switch (i.e., high side active switch) T 1  selectively operable to electrically couple the inductor L out  to a voltage input terminal VIN. A second device T 2  electrically couples the output inductor L out  to a ground GND which is in turn electrically coupled to a ground or common input terminal VIN COM and a ground or common output terminal VOUT COM. 
     As illustrated, the power converter  100  may advantageously take the form of a synchronous buck converter, operable to lower a DC voltage. Where implemented as a synchronous buck converter, the second device T 2  takes the form of a second active switch (i.e., high side active switch), selectively operable to electrically couple the output inductor Lout to ground GND. The power converter  100  may take forms other than a synchronous buck converter, for example a buck converter where the second device takes the form of a passive device, such as a diode (not shown). 
     The switches T 1 , T 2  may take a variety of forms suitable for handling expected currents, voltages and/or power. For example, the switches T 1 , T 2  make take the form of an active device, such as one or more metal oxide semiconductor field effect transistors (MOSFETs). As illustrated in the Figures, the first or high side switch T 1  may take the form of P-Channel MOSFET, while the second or low side switch T 2  make take the form of an N-Channel MOSFET. The output inductor Lout may be coupled via a node  102  to the drains D 1 , D 2  of the MOSFET switches T 1 , T 2  respectively. The power converter  100  may employ other types of switches, for example insulated gate bipolar transistors (IGBTs). While only one respective MOSFET is illustrated, each of the first and/or second switches T 1 , T 2  may include two or more transistors electrically coupled in parallel. 
     The power converter  100  may include an output capacitor Cout electrically coupled between ground GND and a node  104  between the output inductor Lout and the output terminal +VOUT. Output capacitor Cout may smooth the output supplied to the output terminal +VOUT. 
     On an input side, the power converter  100  may include an auxiliary power supply and voltage reference generation block  106 , an over voltage/under voltage monitor block  108  and/or an “in rush” current control block  110 . 
     The auxiliary power supply and voltage reference generation block  106  implements a house keeping supply generation function, amplifier bias generation function and precision reference generation function, resulting in a positive supply voltage or potential VCC, a negative supply voltage or potential or ground VSS, and a precision reference voltage or potential VREF. The structure and operation of the auxiliary power supply and voltage reference generation block  106  can take any existing form, and is not a subject of this application so is not described in further detail. 
     The over voltage/under voltage monitor block  108  monitors instances of over voltage and/or under voltage conditions, supplying a control signal via a control line (not called out in  FIG. 1 ) to the “in rush” current control block  110  as needed. The over voltage/under voltage monitor block  108  or other components may be triggered via an enable signal via an enable input terminal ENABLE. The “inrush” current control block  110  controls “inrush” current, directly limiting current to input capacitor(s) Cin, reducing electrical stresses on the power converter  100  and any system into which the power converter  100  is incorporated. Power converters  100  typically employ large internal bulk filter capacitors to filter the input power to reduce noise conducted out of the power converter  100 , back upstream to the source of the input power. The input capacitor Cin is electrically coupled between ground GND and a node  111  between the “inrush” current control block  110  and the first active switch T 1 . The “inrush” current control block  110  is configured to control the “inrush” current that flows to the input capacitor, particularly at initial application of the input voltage or potential VIN. 
     The structure and operation of the over voltage/under voltage monitor block  108 , the “inrush” current control block  110 , and the input capacitor(s) Cin may take any existing form, and are not subjects of this application so are not described in further detail. 
     Control of the converter circuit (e.g., synchronous buck converter) is realized via a number of components or assemblies, represented in  FIGS. 1 and 2  as blocks. 
     The power converter  100  includes a synchronous gate timing drive control and pulse width modulation (PWM) block  112 . The synchronous gate timing drive control and pulse width modulation block  112  generates gate control signals to control the switches T 1 , T 2 , for example via amplifiers U 1 , U 2 , respectively. The synchronous gate timing drive control and pulse width modulation block  112  may optionally receive a share signal via a share input terminal SHARE from one or more other power converters, for example when electrically coupled to a common load for current sharing operation. The structure and operation of the a synchronous gate timing drive control and pulse width modulation (PWM) block  112  can take any existing form, and is not a subject of this application, so is not described in further detail. 
     The power converter  100  includes an oscillator ramp generation block  114 , also interchangeably referred to herein and in the claims as oscillator or oscillator circuit  114 . The oscillator ramp generation block  114  generates an oscillating ramp signal and provides the oscillating ramp signal to the synchronous gate timing drive control and pulse width modulation block  112 . Advantageously, the oscillator ramp generation block  114  may be selectively synchronized to an external source over a wide frequency range. The oscillator ramp generation block  114  may receive a synchronization signal via a synchronization input terminal SYNC IN, to synchronize operation with one or more other power converters or other devices or systems, for example a clock of a system in which power converter  100  is installed. Such synchronization may advantageously reduce overall system noise. The oscillator ramp generation block  114  may advantageously take a form that provides for external slope modulation of the ramp signal, a wider temperature range and/or an ultra wide synchronous frequency range as compared to existing oscillator ramp generation circuits. The oscillator ramp generation block  114  may additionally, or alternatively, take a form that advantageously employs less complex types of components and/or is less expensive to produce as compared to existing oscillator ramp generation circuits. The structure and operation of exemplary embodiments of the oscillator ramp generation block  114  are described in detail herein with reference to  FIGS. 2-7 . 
     At a high level, the power converter  100  utilizes an inner current control loop and an outer voltage control loop. The inner current control loop is implemented via a current sense block  116 , a current limiting/current sharing (CL/CS) resistor network  118 , a 1-D (one minus duty cycle) compensation block  120  and a current control amplifier  122 . The outer voltage control loop is implemented by a voltage sense resistor divider network  124  (e.g., resistor Rfb coupled between voltage output terminal +VOUT and sense terminal SENSE, divider resistors Rd, Rc, and trim resistors Rb, Ra coupled to trim terminals TRIMB, TRIMA, respectively) and a voltage error amplifier  126  which feeds the CL/CS resistor network  118  to ultimately control the output voltage of the power converter  100 . 
     With respect to the inner current control loop, the current sense block  116  implements current sensing over a portion of a cycle of the power converter  100 , for example over the ON or CLOSED portion of one of the switches T 1 , T 2 . The current sense block  116  provides a signal to the CL/CS resistor divider network  118  to control the current control amplifier  122 , which signal is indicative of the sensed current. For example, the current sense block  116  may sense current over each portion of a cycle during which portion the low side switch T 2  is ON or CLOSED (i.e., conducting), electrically coupling the output inductor L out  to ground GND, while neglecting those portions of the cycle when the low side switch T 2  is OFF or OPEN. 
     Where the output current of the synchronous buck converter circuit in the power converter  100  is sensed at the low side switch (e.g., MOSFET synchronous switch) T 2 , the average of this sensed current is equal to Io*(1-D), where D is defined as the duty cycle of the high side switch (e.g., MOSFET) T 1 . Since this signal is dependent on the duty cycle and negative in value, a compensation signal that is a direct function of the duty cycle is scaled via the 1-D compensation block  120 , and summed with the sensed current signal by the CL/CS resistor network  118 . The resultant signal is optionally level shifted in the CL/CS resistor network  118  to create a level shifted compensated signal. The level shifted compensated signal may then be averaged by the current control amplifier  122 , and the averaged signal used to control the output current of the power converter  100 . 
     The current control amplifier  122  generates control signals based at least on the level shifted compensated signals from the CL/CS resistor divider network  117  to control the synchronous gate timing drive control and pulse width modulation block  112 . 
     With respect to the inner current control loop, the voltage sense resistor network  124  (e.g., resistor Rfb coupled between voltage output terminal +VOUT and sense terminal SENSE, divider resistors Rd, Rc, and trim resistors Rb, Ra coupled to trim terminals TRIMB, TRIMA, respectively) senses voltage or potential at the output terminal +VOUT with respect to the ground terminal VOUT COM. The voltage sense resistor network  124  supplies a signal indicative of the sensed voltage or potential to the voltage sense amplifier  126 . The voltage sense amplifier  126  generates a voltage error signal which indicates a difference between the sensed voltage or potential and a reference voltage or potential. Hence, the voltage sense amplifier  126  is interchangeably referred to herein and in the claims as voltage error amplifier  126 . The voltage error amplifier  126  provides the voltage error signal to the current control amplifier  122  via the CL/CS resistor divider network  118 , for use in generating the control signals supplied to the synchronous gate timing drive control and pulse width modulation block  112  to control output voltage or potential of the power converter  100 . 
     The power converter  100  may optionally include a soft start control block  128 . The soft start control block  128  may receive the precision voltage reference signal VREF from the auxiliary power supply and voltage reference generation block  106 . The soft start control block  128  may control various soft start characteristics of the power converter  100 , for example soft-start time, current limit thresholds, current limit on-time and output voltage or potential level at which control is handed over to a main control loop. The soft start control block  128  may, for example, provide a progressively increasing pulse width, forming a startup voltage ramp which is proportional to a level of a supply voltage VCC, for instance without the need of an external capacitor. The structure and operation of the soft start control block  128  can take any existing form, and is not a subject of this application so is not described in further detail. 
     The topology illustrated in and described with reference to  FIG. 1  is illustrative of only one of the many possible converter topologies which may employ the oscillator ramp generation block  114  described herein. For example, the oscillator ramp generation block  114  may be employed in power converters which use a different converter circuit topology, for instance boost converter, buck converter or fly-back converter topologies. Also for example, the oscillator ramp generation block  114  may be employed in power converters which use a different control topology, for instance a control topology that senses output current over an entire cycle of the waveform without the need for compensation, or which senses current at the high side active switch. Thus, this application, and in particular the claims, should not be limited to the specific topology illustrated in and discussed with reference to  FIG. 1  unless expressly stated therein. 
       FIG. 2  shows the oscillator  114 , according to one illustrated embodiment. 
     The oscillator  114  includes a ramp timing resistor Rramp, timing capacitor Ct, comparator  202 , input voltage divider  204  and reset switch S 1 . The oscillator  114  may include a buffer B 1  to buffer an output signal VRAMP to provide drive capability. The input voltage divider  204  is formed of two resistors Rdiv1, Rdiv2 coupled between the input voltage VSW and a ground GND. The input voltage divider  204  produces a signal indicative of, or proportional to, the input voltage VSW. The comparator  202  is coupled to receive the signal indicative of the input voltage VSW from the input voltage divider  204 . The comparator  202  is also coupled to receive a signal from the timing capacitor Ct indicative of a voltage or electrical potential thereof. The comparator  202  is further coupled a supply voltage or potential VCC and ground reference or potential GND. The oscillator  114  receives an input voltage or potential VSW at an input voltage input pin, terminal or node  206 . The oscillator  114  is guaranteed to start with application of the input voltage or potential VSW at the input voltage input pin, terminal or node  206 . 
     A basic timing sequence of the oscillator  114  includes charging of timing capacitor Ct toward the input voltage or potential VSW. Once the voltage of the timing capacitor Ct reaches the voltage of the input voltage divider  204 , the comparator  202  turns ON the reset switch S 1 , discharging the timing capacitor Ct. The basic timing sequence repeats at a periodic rate or natural or self oscillating frequency of the oscillator  114 . 
     The oscillator  114  also includes synchronization circuitry including a synchronization capacitor Csync, synchronization resistors Rsync 1 , Rsync 2 , and steering logic  208 . The steering logic may include AND circuitry  210  to implement an AND function or operation and OR circuitry  212  to implement a OR function or operation. The synchronization circuitry may receive a synchronization input signal Sync In at a synchronization signal input pin, terminal or node  214 . A first one of the synchronization resistor Rsync  1  is coupled between the input voltage input pin, terminal or node  206  and the synchronization input pin, terminal or node  214 . A second one of the synchronization resistor Rsync  2  is between ground and a node  216  between the synchronization capacitor Csync and part of the OR circuitry  212  steering logic  208 . 
     The synchronization input signal Sync In causes the oscillator  114  to synchronize a frequency of the oscillator  114  to a pulse frequency either below or above the self oscillating frequency of the oscillator  114  without synchronization. The oscillator  114  is capable of locking synchronization, for example with a pulse frequency up to 1 Megahertz. Thus, synchronization signal affects the basic timing sequence of the oscillator. 
     In particular, the AND circuitry  210  and OR circuitry  212  are coupled to cause an early discharge of the timing capacitor Ct in response to the synchronization signal indicating that a higher frequency is required to synchronize (e.g., rising pulse edge), and to delay discharge of the timing capacitor Ct in response to the synchronization signal indicating that a lower frequency is required to synchronize (e.g., logic low). The logic is reflected in Table A, below. 
     
       
         
               
               
               
               
             
           
               
                 TABLE A 
               
               
                   
               
               
                   
                 VCT = 
                 Discharge 
                   
               
               
                 Sync In 
                 VSW? 
                 Ct? 
                 Comment 
               
               
                   
               
             
             
               
                 0 
                 0 
                 0 
                 Not ready for discharge. 
               
               
                 0 
                 1 
                 0 
                 Delay discharge to decrease 
               
               
                   
                   
                   
                 frequency 
               
               
                 ↑ Rising 
                 0 
                 1 
                 Early discharge to increase frequency 
               
               
                 1 
                 1 
                 1 
                 On time discharge, maintain frequency 
               
               
                   
               
             
          
         
       
     
       FIG. 3  shows the oscillator  114 , according to one illustrated embodiment. As illustrated, the oscillator  114  may employ less complex, and less costly discrete semiconductor components, chips resistors and capacitors, than employed by existing oscillators. 
     A capacitor C 52  functions as the timing capacitor Ct ( FIG. 2 ) that is charged through series resistors R 54  and R 54 A which are equivalent to the ramp timing resistance Rramp. The voltage on capacitor C 52  initially charges from ground, exponentially toward the input voltage or potential VSW. The comparator  202  ( FIG. 2 ) is formed by a differential pair of transistors (i.e., first comparator transistor Q 51  and second comparator transistor Q 52 ), with a base of the second comparator transistor Q 52  acting as the non-inverting input and a base of the first comparator transistor Q 51  acting as the inverting input. Emitters of the first and the second comparator transistors Q 51 , Q 52  are coupled to a voltage or potential supply VCC via voltage supply resistor R 58 A and comparator resistor R 53 . The input voltage divider  204  ( FIG. 2 ) is formed by a pair of resistors R 51 , R 52 , which receive the input voltage or potential VSW via resistor R 50 , and which set a comparison threshold on the base of the first comparator transistor Q 51 . The basic timing of the oscillator is governed by the following equations:
 
 Ts=R*C*ln [I−F/R ( t )− F] 
 
 Fs= 1 /Ts  
 
     where: 
     I=initial capacitor voltage 
     F=final capacitor voltage 
     R(t)=comparator threshold. 
     If: 
     F=VSW 
     R(t)=K·VSW 
     I=0 
     Ts=R*C*ln [0−VSW/K·VSW−VSW] 
     Ts=R*C*ln [1/1−K] 
     Thus, frequency is not dependent on input voltage VSW. 
     The timing sequence for the oscillator  114  starts at application of the input voltage or potential VSW, with the timing capacitor C 52  charging up to the comparator threshold, at which point the second comparator transistor Q 52  turns OFF and the first transistor Q 51  turns ON and starts conducting current. This current generates a voltage across a resistor R 55  large enough to charge a capacitor C 51  via a control transistor Q 50  and turn ON a reset or discharge controlling transistor Q 54 . The current used to charge capacitor C 51  flows through the collector of the control transistor Q 50  to develop a voltage across diode CR 53  and control transistor Q 50 . This reduces the threshold voltage at the base terminal of transistor Q 51  to enhance the transition. The reset or discharge transistor Q 54  functions as a voltage clamp that discharges the capacitor C 52  to close to zero volts, as this occurs, a discharge termination transistor Q 55  removes any residual charge stored at the base node of the control transistor Q 50 . 
     A diode CR 50  level shifts resistor R 55  up one diode junction, reducing the voltage across resistor R 55  that is required to turn ON the control transistor Q 50 , and improving the speed of the overall circuit. Once capacitor C 51  discharges through resistor R 56  and reset transistor Q 54  via a reset resistor R 57 , the reset transistor Q 54  turns OFF, allowing the capacitor C 52  to charge again, thereby starting a new timing sequence or cycle. 
     The buffer amplifier B 1  is implemented an emitter follower buffer amplifier by a buffer transistor Q 53  and buffer resistor R 58 , to provide a low source impedance driver capable of driving next stage circuitry. The buffer transistor Q 53  is coupled between the buffer transistor Q 53  and a ground. The buffer resistor R 58  is coupled to a supply voltage or potential VCC and supply voltage resistor R 58 A, and a ground via a capacitor C 53 . 
     The oscillator circuit  114  implements a function whereby a slope of the voltage across capacitor C 52  is modulated proportional to a change in the input voltage or potential VSW. In power supply converters, this type of feature is typically referred to as voltage feed-forward, and substantially improves the line regulation performance of the power converter and reduce changes in the output voltage or potential as a result of changes in the input voltage or potential VSW. A power converter without an oscillator that incorporates voltage feed-forward has degraded input line regulation performance since a change in input voltage or potential must propagate to the output of the converter before the feedback control loop can compensate for the input voltage change. 
     The desired functionality is implemented in the oscillator  114  by making the voltage threshold of the comparator (i.e., Q 51 , Q 52 ) proportional to the input voltage or potential VSW by tying the resistor R 51  of the voltage resistor divider to the input voltage or potential VSW of the power converter. A temperature compensation resistor R 51 A provides temperature compensation to the VSW derived voltage or potential. 
     A logic network comprised of synchronization logic resistors R 59 , R 59 A, synchronization logic diodes CR 52 , CR 51  and synchronization logic capacitor C 50  perform two functions. First, in the case where a frequency of the synchronization signal Sync In is higher than a natural or self frequency of the oscillator circuit  114 , a rising pulse at the synchronize input pin, terminal or node  214  will capacitive couple enough charge through synchronization logic capacitor C 50  and first synchronization logic diode CR 51  to turn on the control transistor Q 50  earlier than might be determined by the timing of the capacitor C 52 . This will terminate the timing charge sequence of the capacitor C 52  early, effectively increasing the oscillator frequency of operation. Second, in the case where a frequency of the synchronization signal Sync In is lower than the natural frequency of the oscillator circuit, a low voltage of the pulse at the synchronize input pin, terminal or node  214  through the second synchronization logic diode CR 52  will hold off or delay the reset (i.e., discharging) of the capacitor C 52 , effectively lowering the oscillator frequency of operation. 
     The illustrated oscillator circuit  114  may be capable of operation up to, for example 1 MHz. Such high speed operation is accomplished via use of a differential transistor pair topology, optimized to minimize all voltage transitions and advantageously employing a minimum number of components. 
     The above described oscillator designs implement slope modulation, self starting operation, synchronization, wide frequency and wide temperature range of operation. The oscillator design described herein incorporates all these features without the use of complex integrated circuits. Rather, the oscillator design implements all of the desired features with simple, low cost devices while maintaining the desired performance over wide temperature and operating conditions. Designs utilizing complex integrated circuits tend to be more costly, require substantially more parts, and do not appear capable of operating over as wide a frequency range and temperature range. 
       FIG. 4  shows a flow diagram of a method  400  of operating the oscillator  114  of  FIGS. 1-3 , according to one illustrated embodiment. 
     At  402 , an oscillator circuit  114  receives an input voltage or potential signal VSW at an input voltage input pin, terminal or node  206 . The input voltage or potential signal may be the input voltage or potential supplied to the power converter  100  ( FIG. 1 ) from some upstream component. For example, the input voltage or potential may be supplied to the power converter from a rectifier, a DC/DC converter, an isolating converter stage, and/or a DC electrical power storage device such as an array of chemical battery cells or ultra-capacitors. 
     At  404 , the oscillator circuit  114  receives a synchronizing signal Sync In at a synchronization signal input pin, terminal or node  214 . For example, a synchronizing signal Sync In may come from another power converter coupled in parallel with the power converter  100  ( FIG. 1 ) to a common load and operating as either a master or a slave in a current sharing arrangement to supply current to the common load. 
     At  406 , a ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) is charged through a ramp timing resistance Rramp, R 54 , R 54 A using the voltage or potential input VSW. 
     At  408 , a voltage divider  204  ( FIG. 2 ), R 51 , R 52  ( FIG. 3 ) produces a discharge trigger reference voltage. The discharge trigger reference voltage may be proportional to a voltage or potential of the input voltage signal VSW. 
     At  410 , logic circuitry  208  ( FIG. 2 ) determines if a state of a synchronization signal Sync In indicates that frequency needs to be increased, decreased or stay the same. As explained above, a logic network ( FIG. 3 ) comprised of synchronization logic resistors R 59 , R 59 A, synchronization logic diodes CR 52 , CR 51  and synchronization logic capacitor C 50  may advantageously implement the logic without the need for complicated integrated circuits. 
     If the synchronization signal Sync In indicates that the frequency needs to be increased (e.g., rising edge), then discharging circuitry fully discharges the ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) at  414 . The discharging circuitry may include switch S 1  ( FIG. 2 ) coupled to ground or reset transistor Q 54  ( FIG. 3 ) coupled to ground. 
     If the synchronization signal Sync In indicates that the frequency needs to be decreased (e.g., low logic state), then charging of the ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) continues at  416 . As explained above, such be indicated by a falling pulse applied to the control transistor Q 50  ( FIG. 3 ) via synchronization logic diode CR 52 . 
     If the synchronization signal Sync In indicates that the frequency should remain the same (e.g., high logic state), then control passes to  412 . 
     At  412 , a comparator  202  determines whether a voltage of the ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) is equal to the discharge trigger reference voltage. As noted, a differential pair of transistors Q 51 , Q 52  ( FIG. 3 ) may be advantageously employed to perform the comparison. 
     If the voltage of the ramp timing capacitor C 52  is equal to the discharge trigger reference voltage, then discharging circuitry fully discharges the ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) at  414 . Otherwise, charging of the ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) continues at  416 . 
     At  424 , a buffer amplifier B 1  buffers an output voltage signal VRAMP. As illustrated in  FIG. 3 , the output voltage signal VRAMP may be buffered by an emitter follower buffer amplifier having a discrete buffer transistor Q 53  and buffer resistor R 58 . 
     The method  400  may repeat while the power converter  100  is operational, the oscillation circuit  114  continually generating, producing or supplying the oscillation ramp signal VRAMP. 
       FIG. 5  shows a method  500  of operating the oscillator  114  of  FIGS. 1-3 , according to one illustrated embodiment. The method  500  may be implemented as part of performing the method  400  of  FIG. 4 . In particular, the method  500  specifies how the voltage at the ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) may be compared to the discharge trigger reference voltage. 
     At  502 , a discharge trigger reference voltage is supplied to a first comparator transistor Q 51  of a differential pair of transistors. For example, the supply discharge trigger reference voltage from an input voltage divider R 51 , R 52  ( FIG. 3 ) may be supplied to a base of the first comparator transistor Q 51 , which has an emitter commonly coupled with an emitter of a second comparator transistor Q 52 . 
     At  504 , a voltage across ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) is supplied to the second comparator transistor Q 52  of the differential pair of transistors. For example, the ramp timing capacitor voltage or potential may be supplied to a base of the second comparator transistor Q 52 . 
       FIG. 6  shows a method  600  of operating the oscillator  114  of  FIGS. 1-3 , according to one illustrated embodiment. The method  600  may be implemented as part of performing the method  400  of  FIG. 4 . The method  600  may determine whether to trigger discharging of the ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) when the value of the ramp timing capacitor voltage is equal to the discharge trigger reference voltage. 
     At  602 , a synchronization signal Sync In is supplied to a logic network (e.g., synchronization logic resistors R 59 , R 59 A, synchronization logic diodes CR 52 , CR 51  and synchronization logic capacitor C 50  of  FIG. 3 ). 
     At  604 , discharging of ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) is triggered via the first synchronization diode CR 51  in response to a state of the synchronization signal indicating that frequency needs to be increased (e.g., rising edge of pulse of synchronization signal Sync In). 
     At  606 , discharging of ramp timing capacitor Ct ( FIG. 2 ), C 52  ( FIG. 3 ) is suppressed via the second synchronization diode CR 52  in response to a state of the synchronization signal indicating that frequency needs to be decreased (e.g., falling edge of pulse of synchronization signal Sync In). Such may be suppressed even though the normal discharging triggering condition of ramp timing capacitor voltage being equal to the discharge trigger reference voltage is satisfied. 
       FIG. 7  shows a method  700  of operating the oscillator  114  of  FIGS. 1-3 , according to one illustrated embodiment. The method  700  may be implemented as part of performing the method  400  of  FIG. 4 . For example, the method  700  may be employed to produce, generate or supply a discharge trigger reference voltage that is proportional to a voltage or potential of the input voltage signal. 
     At  702 , an input voltage resistor divider network  204  ( FIG. 2 ) R 51 , R 52  ( FIG. 3 ) divides an input voltage signal VSW to produce a proportional signal. 
     At  704 , a temperature compensation resistor R 51 A compensates the divided input voltage signal for variations in temperature. 
     The described oscillator circuit  114  may provide an oscillating ramp signal where a slope of the ramp is modulated in response to an externally applied signal. Such may have linearity over a wide range of operating frequencies and/or temperatures. The described oscillator circuit  114  may allow synchronization with an externally applied synchronization signal. Such may be advantageously implemented using relatively simple and inexpensive components. 
     While described above in the environment of a power converter, and in particular a switch mode DC/DC synchronous buck power converter, the oscillator circuitry may be advantageously employed in a large variety of other environments. Such may include other types of DC/DC power converters (e.g., boost, buck-boost, flyback), whether isolated or non-isolated. Such may also include other types of power converters, including inverters (DC/AC). Such may also include many other types of non-power converter environments, which employ an oscillating ramp signal, for example to generate, produce or supply PWM signals to various types of devices. 
     For example, the oscillator and voltage ramp generator may be employed in a simple buck converter topology. 
     A converter switch of the buck converter may be operable to selectively couple an inductor of the buck converter to the input voltage VIN and a ground reference via a ground diode. 
     The converter switch may be controlled by the output of a pulse width modulator ramp comparator. The pulse width modulator ramp comparator receives a ramp voltage V Ramp  from the oscillator and voltage ramp generator on a positive or non-inverting input pin. The pulse width modulator ramp comparator receives a control signal V control  on a negative or inverting pin, from an output of an error voltage integrator control amplifier. 
     The error voltage integrator control amplifier receives a reference voltage VREF on a positive or non-inverting pin. The error voltage integrator control amplifier receives a signal indicative of an output voltage on a negative or inverting pin thereof via an integration resistor R I . An integrator feedback capacitor C F  and resistor R F  feeds the output (i.e., V control ) of the error voltage integrator control amplifier back to the negative pin thereof. 
     Thus, the difference between the reference voltage VREF and the output voltage VOUT is applied to the error voltage integrator control amplifier. This circuit comprised of integration resistor R 1 , feedback resistor RF, feedback capacitor CF and the amplifier creates the time averaged control voltage VControl, which represents the drive energy required to maintain a desired output voltage VOUT. The control voltage VControl is applied along with the ramp voltage VRamp from the oscillator and voltage ramp generator to the pulse width modulator ramp comparator to create a pulse width modulation signal used to control a state of converter switch (e.g., transistor). The pulsed ON time of converter switch provides an average current that is translated to a continuous output voltage VOUT via an output filter circuit comprised of the diode, the inductor and a load capacitor C Load . The output node VOUT is used to supply power to loads requiring a fixed DC output voltage. 
     With the output voltage VOUT maintaining a constant DC voltage and for any given DC load connected to the output terminal VOUT, it is desirable to maintain a relatively constant control signal V control  independent of the applied input voltage VIN. 
     The oscillator and voltage ramp generator samples the input voltage VIN to modulate the rate of the ramp voltage (dV/dT) proportionally to the input voltage VIN, while at the same time maintaining a fixed frequency. The increased rate (dV/dT) of the ramp voltage VRamp causes the output to reach the control V control  comparison more quickly, forcing the pulse width modulator ramp comparator to reduce an average ON time of the converter switch at the fixed PWM frequency. 
     The specific values, such as specific voltages or potentials, used herein are purely illustrative, and are not meant to be in anyway limiting on the scope. Likewise, the arrangements and topologies are merely illustrative and other arrangements and topologies may be employed where consistent with the teachings herein. While specific circuit structures are disclosed, other arrangements that achieve similar functionality may be employed. The terms switched mode and switch mode are used interchangeable herein and in the claims. 
     The methods illustrated and described herein may include additional acts and/or may omit some acts. The methods illustrated and described herein may perform the acts in a different order. Some of the acts may be performed sequentially, while some acts may be performed concurrently with other acts. Some acts may be merged into a single act through the use of appropriate circuitry. 
     The various embodiments described above can be combined to provide further embodiments. All of the U.S. patents, U.S. patent application publications, U.S. patent applications, foreign patents, foreign patent applications and non-patent publications referred to in this specification and/or listed in the Application Data Sheet, including but not limited to commonly assigned U.S. patent applications: 
     Ser. No. 13/185,142, titled “POWER CONVERTER APPARATUS AND METHOD WITH COMPENSATION FOR LIGHT LOAD CONDITIONS”; 
     Ser. No. 13/185,217, titled “SELF SYNCHRONIZING POWER CONVERTER APPARATUS AND METHOD SUITABLE FOR AUXILIARY BIAS FOR DYNAMIC LOAD APPLICATIONS”; 
     Ser. No. 13/185,210, titled “INPUT CONTROL APPARATUS AND METHOD WITH INRUSH CURRENT, UNDER AND OVER VOLTAGE HANDLING”; 
     Ser. No. 13/185,172, titled “POWER CONVERTER APPARATUS AND METHOD WITH COMPENSATION FOR CURRENT LIMIT/CURRENT SHARE OPERATION”; and 
     Ser. No. 61/508,937, titled “POWER CONVERTER APPARATUS AND METHODS”; 
     all filed on Jul. 18, 2011, are incorporated herein by reference, in their entirety. Aspects of the embodiments can be modified, if necessary to employ concepts of the various patents, applications and publications to provide yet further embodiments. 
     These and other changes can be made to the embodiments in light of the above-detailed description. In general, in the following claims, the terms used should not be construed to limit the claims to the specific embodiments disclosed in the specification and the claims, but should be construed to include all possible embodiments along with the full scope of equivalents to which such claims are entitled. Accordingly, the claims are not limited by the disclosure.