Abstract:
A circuit for a switch mode power supply is presented. The circuit comprises a transient detection portion adapted to delay an analogue error signal (Vdiff) derived from the output voltage (Vout) of the switch mode power supply and to detect whether the difference between the output voltage and the delayed analogue error signal (Vdel) is within a predetermined range.

Description:
[0001]    This application claims the priority under 35 U.S.C. §119 of European patent application no. 09179269.7, filed on Dec. 15, 2009, the contents of which are incorporated by reference herein. 
         [0002]    This invention relates to switch-mode power supplies (SMPS), and more particularly to a circuit for a SMPS. 
       BACKGROUND OF THE INVENTION 
       [0003]    Switch-mode power supplies (SMPS) used in consumer electronics, portable applications, and computers, are required to meet stringent voltage regulation requirements using a cost-effective implementation occupying a small volume. The regulation is usually achieved with on-chip integrated controllers which typically utilize voltage mode pulse-width modulation (PWM). 
         [0004]    When a load change occurs, it is the task of the supply regulator to keep the supply voltage within a well-defined range in order to prevent overstress or performance degradation. 
         [0005]    A fast reaction to such load changes is required to prevent too much deviation from the nominal supply-value. The traditional voltage mode PID control is not very suited for this, whether it is implemented in an analogue or digital way, mainly because the regulation parameter, usually the duty-cycle of the PWM control signal, is only updated once in a switching cycle. Thus, even with large gain-settings, it can take up to a switching period before a reaction is due. 
         [0006]    A fast load change, in either direction, is often referred to as “transient”. Methods to recover quickly from such transients, called “transient improvement”, typically aim both to minimize the over- or under-shoot as well as to shorten the recovery time. Recovery time is defined as the time needed from the start of the transient to the point where the output voltage is back within the required steady-state regulation band. 
         [0007]    Many methods of transient improvement have been proposed, such as Active Transient Response by Primarion and Non Linear Response by Zilker, These have recently gained acceptance due to the availability, and commercially affordability, of digital techniques. 
         [0008]    Such methods try to react as soon as possible to a load transient and act accordingly. 
         [0009]    Usually, an error signal is obtained and processed for analysis before the decision is made a real transient occurred. During this processing, however, valuable time is wasted, leading to an already large deviation of the output voltage from its reference value. The sooner corrective action can be taken, i.e. starting to transfer maximum possible energy to the output in the case of a light-to-heavy load change, or, stopping immediately any energy transfer in the case of a heavy-to-light load change, the smaller the resulting under- or over-shoot, respectively, will be. 
         [0010]    Thus, for a good transient response, it is important to have a transient detection as fast as possible, whilst also being reliable and distinctive. 
         [0011]    When corrective action has been taken (either maximum energy transfer or none at all), the output voltage (V out ) will eventually reach its maximum deviation (called the valley or peak, respectively). For many improvement methods, it is important to precisely determine the moment the valley or peak occurs. In other words, it is desired to accurately determine the moment when dV out /dt=0. Any substantial deviation or error in such detection will render the response non-ideal. It should be noted here that such a peak or valley occurs in a buck convertor (a SMPS which is a step-down DC to DC converter) on the moment when the current through the inductor becomes equal to the load current drawn from the convertor output. 
         [0012]    A complication here is the fact that the measured V out  is not the real voltage on the intrinsic output capacitor C out , due to the parasitic series resistance ESR (R esr ) of the capacitor. It can be easily derived the voltage on the intrinsic capacitor will lag by a time t esr =R esr ×C out , for which a correction can be made when determining the peak/valley by observing V out . 
         [0013]    Shortly after a load change occurs, a second change may occur, in the same or opposite direction. This is commonly referred to as “multi load step”. Many methods of transient improvement lack the possibility to detect such consecutive transients, while still handling the first one, possibly leading to unwanted large output voltage excursions. 
         [0014]    Among the most commonly used solutions in digital implementations for obtaining the error signal, being the difference between the voltage set-point V ref  and the output voltage V out , the following methods are known: 
         [0015]    1) A low resolution Flash ADC, converting only the deviation of V out . This is small and efficient, but suffers from a limited semi-fixed conversion range (which only changes when the set-point V ref  changes). 
         [0016]    2) A very fast high-resolution ADC, covering the total V out  range. This has the drawback of requiring a large silicon area and is power hungry. 
         [0017]    3) A so-called Track-and-Detect ADC which is small and effective in transient detection, but still requires error tracking in order to determine the peak or valley. 
         [0018]    The most common way of digitising the error signal is by using a differential amplifier followed by a fast Flash ADC, as depicted in  FIG. 1 . 
         [0019]    The differential amplifier  10  produces the differential voltage of the two terminals V out     —     p  and V out     —     n , between which the generated output voltage V out  is regulated. V out  is connected to all positive inputs ‘in’ of first to Nth comparators  12 . The reference inputs ‘ref’ of the comparators  12  each have a different reference level, both positive and negative, and not necessarily equidistant, with respect to the reference voltage Vref, in essence the set-point. For each comparator whose input ‘in’ sees a higher voltage than its reference input ‘ref’, the output will be high. Thus, the generated output of the Flash ADC will behave like a thermometer code, which may be subsequently encoded into a form of error code. 
         [0020]    The number N of comparators  12 , in combination with the reference levels, determines the accuracy and the useful range of the error measurement. For an accurate peak-/valley-detection a small difference between the reference levels is preferred. If the peak or valley always occurs in the same region, this solution may be adequate, but this is not usually the case and thus a large number of comparators is generally needed, making this known solution unsuitable for many applications. 
         [0021]    Another way of addressing the accuracy issue is to apply a full-fledged, fast, high-resolution ADC (as shown in  FIG. 2 ), to digitise the absolute value of V out  with the required number of bits. Such an ADC, however, usually consumes a large amount of current and occupies a relatively large area in silicon. Moreover, a transient can only be detected by processing the error signal, unless two extra comparators are added to perform this function, but with the same multi load step detection problem as in the previous solution. 
         [0022]    A third option is the so-called Track-and-Detect ADC, as depicted in  FIG. 3 . 
         [0023]    Here, two (static) comparators  30  are used for steady-state regulation and two other (dynamic) comparators  32  are used for transient- and peak-/valley-detection, thus requiring only a very small silicon area. However, this approach lacks the possibility to detect a transient faster than via error processing. It will be appreciated that this can be solved by adding two dynamic comparators, with their reference levels forming a somewhat larger window than the peak/valley detection window, to retain the transient detection at all times. Nonetheless, the peak-/valley-detection accuracy depends on the achievable detection window of the two dynamic comparators  32 . 
         [0024]    A smaller “virtual” window can be obtained by first amplifying the error signal (Vout−Vref), by means of a differential amplifier  34  (having a gain=1+R/R=2 in the arrangement of  FIG. 3 ), before feeding the output (Vdiff) of the differential amplifier  34  to the dynamic comparator  32  inputs ‘in’. Here, a simple (analogue) way of filtering spikes and noise is provided by using a feedback capacitor  36 . 
         [0025]    Whenever Vdiff hits either side of the peak/valley detection window, defined by the reference levels of dynamic comparators  32 , the logic gate  38  will close switch  40  until the voltage on hold capacitor  42  is within the detection window again, and so on and so forth, making the voltage on hold capacitor  42  effectively follow Vdiff. Thus, a peak or valley can always be detected by observing a change in the sign of the error signal, derived from the dynamic comparator  32  outputs. 
         [0026]    As mentioned above, although the arrangement of  FIG. 3  is effective in transient detection, it still requires error tracking in order to determine a peak or valley. 
       SUMMARY OF THE INVENTION 
       [0027]    According to the invention, there is provided a circuit for a switch mode power supply comprising a transient detection portion adapted to delay an analogue error signal derived from the output voltage of the switch mode power supply and to detect whether the difference between the output voltage and the delayed analogue error signal is within a predetermined range. 
         [0028]    Thus, there is provided an approach to undertaking fast transient detection and accurate peak/valley-detection. Embodiments may be accurate to whatever level the output voltage excursion may reach (within a normal operating range). They may also be capable of detecting another load change whilst an initial change has not yet been fully counteracted. 
         [0029]    The transient detection portion may comprise a resistor-capacitor, RC, combination. An RC combination can have a fairly stable RC product, which can be made easily adaptive in silicon. 
         [0030]    The transient detection portion may be adapted to determine if the delayed analogue error signal is smaller or larger than the analogue error signal. A simple voltage comparison may thus be sufficient to accurately determine the location of a real peak or valley. 
         [0031]    Embodiments may employ the use of an analogue delay line and three comparators, therefore providing a simple solution having reduced power and area requirements. 
         [0032]    According to another aspect of the invention, there is provided a method for regulating the output voltage of a switch mode power supply comprising the steps of: deriving an analogue error signal from the output voltage of the switch mode power supply; delaying the analogue error signal; and detecting whether the difference between the output voltage and the delayed analogue error signal is within a predetermined range. 
         [0033]    Embodiments may be used in loop regulators for Point of Load applications or in other applications where a need exists for fast transient detection and/or accurate peak-/valley-detection. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0034]    Examples of the invention will now be described with reference to the accompanying drawings, in which: 
           [0035]      FIG. 1  illustrates a known Flash ADC arrangement for digitising an error signal; 
           [0036]      FIG. 2  shows a conventional high-resolution ADC; 
           [0037]      FIG. 3  illustrates a conventional arrangement referred to as a Track-and-Detect ADC; 
           [0038]      FIG. 4  shows an integrated circuit arrangement according to an embodiment of the invention; 
           [0039]      FIG. 5  shows typical waveforms of signals of the embodiment of  FIG. 4 ; 
           [0040]      FIG. 6  shows an integrated circuit arrangement according to another embodiment of the invention; 
           [0041]      FIG. 7  is a simplified schematic of a buck convertor employing the embodiment of  FIG. 6 ; 
           [0042]      FIG. 8  shows the simulated behaviour of the combination of a controller, according to the embodiment of  FIG. 6 , and a buck converter having an inductor of 320 nH and an output capacitor of 400 μF with an ESR of 1 mΩ, responding to a 0-30 A load step; and 
           [0043]      FIG. 9  illustrates the improvement in transient response for a 0-30-0 A load current step between a preferred embodiment of the invention, a PID compensator and a state-of-the-art controller for a DC-DC convertor. 
       
    
    
     DETAILED DESCRIPTION OF EMBODIMENTS 
       [0044]    The drawings are not to scale, and some dimensions may have been exaggerated (for example the thickness dimension) to make the drawings show the different components more clearly. 
         [0045]    Referring to  FIG. 4 , it can be seen that the illustrated embodiment is similar to the known circuit of  FIG. 3  except that the comparator arrangement used for transient-and peak-/valley-detection is different. Specifically, the illustrated embodiment comprises an analogue delay line  45  and three fast comparators  46 ,  48  and  50 . 
         [0046]    Similarly to the known arrangement of  FIG. 3 , the embodiment obtains the analogue error signal Vdiff (=Vout−Vref) using a differential amplifier  34 , whose inverting input is connected to Vref and its non-inverting input to Vout. 
         [0047]    A (small) feedback capacitor  36  is connected between the output of amplifier  34  and its inverting input to provide filtering of excessive noise and spikes. 
         [0048]    The (amplified) analogue error signal Vdiff provided by the output of the differential amplifier  34  is connected both to steady-state comparators  30  used for the integrator and differentiator with proportional gain (PID) compensator, as well as to the three fast comparators  46 ,  48  and  50  for the transient- and peak-/valley-detection circuit. The gain factor of the amplifier  34  can be chosen such that a requirement for offset voltage of the comparators can be met easily. 
         [0049]    The reference inputs of the steady-state comparators  30  are connected to fixed levels with respect to Vref, while the reference inputs ‘VTU’ and ‘VTD’ of the fast comparators  48  and  50  in the detection circuit are connected to fixed levels with respect to a dynamic reference ‘Vdel’. The level of this dynamic reference ‘Vdel’ is solely determined by the output of an analogue delay line  45 , having the original analogue error signal Vdiff as its input. 
         [0050]    If a load transient occurs and the excursion of Vdiff crosses either side of the transient detection window, formed by the reference levels ‘VTU’ and ‘VTD’, and the level crossing occurs within the propagation delay time of the delay line, the transient will be detected and indicated by a state change of one of the dynamic comparators  48  and  50 . In this way, the detection circuit detects whether the output voltage is within a predetermined range of the delayed analogue error signal Vdel, wherein the range is set by the reference levels ‘VTU’ and ‘VTD’. In other words, it is detected whether the absolute difference between the output voltage and the delayed analogue error signal Vdel is less than a predetermined threshold value. 
         [0051]    The larger the gradient/differential di/dt of the transient, the sooner it will be detected and the better it can be compensated, whereas slow or shallow transients will not trigger the comparators  48  and  50 . However, such transients can be compensated adequately by the PID compensator alone. 
         [0052]    It will be understood that a small propagation delay of the error amplifier  34  and comparators  48  and  50  will significantly contribute to the achievable transient detection delay and, hence, to the achievable under- or over-shoot. 
         [0053]    Accurate detection of a peak or valley is achieved by observing the fast comparator  46  output. 
         [0054]      FIG. 5  shows typical waveforms of both the error amplifier output Vdiff and the delay line output Vdel (as solid lines), along with the transient detection levels ‘VTU’ and ‘VTD’ (as dash-dotted lines). 
         [0055]    At the bottom of the figure the output levels of comparator  46  and the lower comparator  50  are also illustrated. 
         [0056]    It has been found that a peak or valley is always very close to the crossing point of the error signal Vdiff with its delayed version Vdel, which can be easily detected via the comparator  46  output state. The requirement for a unity gain of the delay line  45  will be apparent, as otherwise the crossing point of Vdiff and Vdel will move away from the optimal point. It will also be appreciated that the propagation delay of the delay line itself may influence the position of the crossing point and so detection may be less accurate for faster curves where the propagation delay of the delay line is significant. 
         [0057]    When considering a practical implementation of the analogue delay line, a simple approach may use a resistor-capacitor (RC) combination. However, an RC-delay does not exhibit unity gain across a range of frequencies, nor an adequate group delay. However, an RC combination can have a fairly stable RC product, which can even be made easily adaptive in silicon. 
         [0058]    In spite of the abovementioned drawbacks, an RC-delay has an advantage which helps to demonstrate the simplicity of embodiments according to the invention. To demonstrate this,  FIG. 6  shows an embodiment wherein the delay line  45  is implemented using an RC combination. The RC combination comprises a resistor  60  and capacitor  62 . 
         [0059]    The RC time-constant of this delay line is chosen to equal the time-constant t esr , determined by the output capacitance and the equivalent series resistance of the output buffer capacitor of the buck converter, such that the voltage V RC  on capacitor  62  will be a reproduction of the virtual voltage on the intrinsic output capacitor. 
         [0060]    In this way, the ‘real’ peak or valley can now be determined by performing peak/valley detection on the real available voltage V RC  on the capacitor  62 . In this regard, it is recalled that the real peak or valley occurs when the current through the inductor is equal to the load current drawn from the output of the buck converter. Thus, precisely at the moment of the real peak or valley, the voltages V out  and V RC  are necessarily equal, since no current will be flowing through the ESR of the output capacitor ‘CO’ or through the resistor  60 , as illustrated in  FIG. 7 . 
         [0061]    Similarly, whenever the load current is larger than the inductor current, V out  will be smaller than V RC  and vice versa. 
         [0062]    Thus, the practical implication of this phenomenon is the fact that a simple comparison of V diff  and V RC  is sufficient to determine the exact location of the real peak or valley. 
         [0063]    Turning now to  FIG. 8 , shown is the simulated behaviour of the combination of a controller, according to the embodiment of  FIG. 6 , and a buck converter having an inductor of 320 nH and an output capacitor of 400 μF with an ESR of 1 mΩ, responding to a 0-30 A load step. 
         [0064]    The transient detection window is +/−25 mV, but effectively +/−12.5 mV due to the gain of two in the differential amplifier. Resistor  60 =4 kΩ, Capacitor  62 =100 pF. The steady-state regulation window is +/−5 mV, so effectively +/−2.5 mV at V ref =1.5V. 
         [0065]    It can be clearly seen that V diff  crosses V RC  exactly at the point where the current through the inductor (I_inductor) equals the current through the load (I_load). 
         [0066]    In addition to the simulation, an experimental prototype of the invention has been built using a field programmable gate array (FPGA) for all logic circuitry and a slightly modified existing printed circuit board (PCB) containing all analogue and mixed-signal circuitry including all components of the buck-converter. 
         [0067]      FIG. 9  shows the improvement in transient response for a 0-30-0 A load current step between the preferred embodiment (labelled “Invention”), the PID compensator only (labelled “PID-only”) and a state-of-the-art controller for a DC-DC convertor (labelled “[ 1 ]”). 
         [0068]    The scale for V out  is 50 mV/div, for I_inductor it is 5 A/div and the time base is 5 μs/div. 
         [0069]    The PID compensator with only six comparators is not optimised for speed, because that would yield a poor steady-state regulation. The response of the state-of-the-art controller for a DC-DC convertor is actually measured using existing silicon of the TEA1321 and the load step duration is made somewhat shorter to prevent too much overlap with the response of the preferred embodiment of the invention. 
         [0070]    As can be seen, the main improvement of the present invention with respect to a state-of-the-art approach is about three times smaller undershoot and faster recovery to within the regulation band. 
         [0071]    A preferred embodiment of the present invention may be equipped with a dual set of PID settings, one slow set used during steady-state (when Vout is within regulation band), the other, fast set, when not in steady-state and not in transient mode, thus enabling faster recovery and almost flat current response. 
         [0072]    For an even better synchronisation of the handing back of the control to the PID controller after a transient recovery the second peak or valley (I_inductor=I_load) may be used, as at that moment both voltage and current are close to the new steady-state, which guarantee a smoother transition to PID mode because of the smaller error. 
         [0073]    The invention can be used in loop regulators for Point of Load applications, or in any other application where a need exists for (fast) transient detection and/or (accurate) peak-/valley-detection. 
         [0074]    Other variations to the disclosed embodiments can be understood and effected by those skilled in the art in practicing the claimed invention, from a study of the drawings, the disclosure, and the appended claims. In the claims, the word “comprising” does not exclude other elements or steps, and the indefinite article “a” or “an” does not exclude a plurality. Any reference signs in the claims should not be construed as limiting the scope.