Abstract:
A sensing circuit for sensing a memory cell, the sensing circuit having a first circuit branch electrically connectable to the memory cell to receive a memory cell current, the first circuit branch having at least one first transistor that, when the first circuit branch is connected to the memory cell, is coupled thereto substantially in a cascode configuration. A bias current generator is operatively associated with the first transistor for forcing a bias current to flow therethrough.

Description:
TECHNICAL FIELD 
     The present invention relates in general to the field of semiconductor memories, particularly but not limitatively to non-volatile memories such as ROMs, EPROMs, EEPROMs and Flash EPROMs or EEPROMs. More specifically, the invention concerns a sensing circuit for sensing the information stored in memory cells, and more particularly to a sensing circuit adapted to the sensing of multi-level memory cells programmable in more than two programming states for storing more than one bit of information. 
     BACKGROUND OF THE INVENTION 
     In non-volatile semiconductor memories such as EPROMs, EEPROMs and Flash EPROMs or EEPROMs, the information is stored as a charge trapped in a floating gate electrode of a floating-gate MOS transistor memory cell. One bit of information can be stored by means of two different charge values, which correspond to two different values or levels of the MOS transistor threshold voltage. 
     In order to sense the single bit of information stored in a memory cell, the latter is biased in a prescribed sense condition and the current sunk by the memory cell is detected, normally by comparison with a reference current. The memory cell sense condition is chosen so that, depending on the charge trapped in the floating gate, the memory cell either conducts a current or does not conduct any current. 
     In recent years the possibility of storing more than one bit of information in each memory cell has been proposed. More than one bit of information can be stored in a memory cell provided that the number of possible charge values trapped in the floating gate electrode is increased. This corresponds to increasing the number of different possible threshold voltage levels of the memory cell. The memory cell is therefore referred to as multi-level, in contrast to the two-level memory cell in which only two threshold voltage levels exist. 
     For example, a four-level memory cell stores two bits of information, while a sixteen-level memory cell stores four bits of information. 
     Memory devices with multi-level memory cells substantially increase the storage capacity per unit area. 
     The introduction of multi-level memory cells has led to the implementation of sensing circuits that attempt to maximally exploit the memory cell operating window. The control gate electrode of the memory cell is thus biased at the maximum potential allowed by the fabrication technology, compatibly with the performance required of the memory device in terms of retention of the stored data. Nowadays, the typical sense potential applied to the memory cell control gate electrode is approximately 6 V. 
     Additionally, the necessity of allocating several different threshold voltage levels, for example four or even sixteen, leads to the maximum distribution thereof between the minimum and maximum threshold voltage levels which can be detected by the sensing circuit. 
     Clearly, for low threshold voltage levels the memory cell sinks relatively high currents, while for threshold voltage levels close to the maximum detectable value the current sunk by the memory cell is very small and becomes zero for the highest threshold voltage level. 
     As a consequence, the biasing conditions of the sensing circuits significantly changes in dependence of the status of the memory cell to be sensed, that is of the memory cell threshold voltage. 
     The changes in the biasing conditions of the sensing circuits may be unacceptable in terms of the different length of the transients, which causes different access times depending on the fact that the accessed memory cell has a low or high threshold voltage level. 
     Additionally, the biasing condition of the memory cell, particularly the bias voltage of the drain electrode of the floating-gate MOS transistor, varies in dependence of the memory cell programming state: for low threshold voltage levels, corresponding to relatively high currents sunk by the memory cell, the drain voltage lowers, while for high threshold voltage levels, corresponding to low currents, the drain voltage rises. These changes in the drain biasing condition may attenuate the signal to be detected in a non-linear way. 
     SUMMARY OF THE INVENTION 
     The disclosed embodiments of the present invention provide a sensing circuit structurally and functionally adapted to overcome the drawbacks of the prior-art circuits. 
     According to a first aspect of the present invention, a sensing circuit for sensing a memory cell is provided. The sensing circuit comprises a first circuit branch electrically connectable to the memory cell so as to be run through by a memory cell current. The first circuit branch includes at least one first transistor which, when the first circuit branch is connected to the memory cell, is coupled thereto substantially in a cascode configuration. A bias current generator is operatively associated with the first transistor for forcing a bias current to flow therethrough. 
     In one embodiment, the bias current generator comprises a first current injector for injecting a first current into a first electrode of the first transistor, and a first current extractor for extracting a second current from a second electrode of the first transistor. 
     Advantageously, the first current extractor comprises a first two-branch current-mirror circuit with a first current-mirror circuit branch run through by a predetermined current, a second current-mirror circuit branch sinking the second current, and an operational amplifier for controlling a conductivity of the second current-mirror circuit branch so as to keep the second current at a prescribed value, fixed by the predetermined current. 
     According to another aspect of the present invention, there is provided a current extractor circuit for extracting a prescribed current from a circuit node. The current extractor circuit comprises a two-branch current-mirror circuit with a first current-mirror circuit branch run through by a predetermined current, a second current-mirror circuit branch for extracting the prescribed current, and an operational amplifier for controlling a conductivity of the second current-mirror circuit branch so as to keep the second current at a prescribed value, fixed by the predetermined current. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     The features and advantages of the present invention will be made apparent by the following detailed description of some embodiments thereof, provided merely by way of non-limiting examples and illustrated in the annexed drawings, wherein: 
     FIG. 1 is a schematic circuit diagram of a sensing circuit according to a preferred embodiment of the present invention; 
     FIG. 2 shows a portion of a sensing circuit according to a first alternative embodiment of the present invention; 
     FIG. 3 shows a portion of a sensing circuit according to a second alternative embodiment of the present invention; 
     FIG. 4 is a circuit diagram of a possible embodiment of a current generator circuit for any one of the sensing circuits of the preceding figures, and 
     FIG. 5 reports in diagrammatic form the result of simulations conducted by the Applicant on the current generator circuit of FIG.  4 . 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     In the following, same reference numerals will be adopted to identify same, similar or corresponding parts in the different embodiments which will be described. 
     FIG. 1 schematically shows a sensing circuit  10  according to a preferred embodiment of the present invention. The sensing circuit  10  will normally be incorporated in a memory device, either stand-alone or embedded in a more complex integrated circuit. More specifically, the sensing circuit  10  will normally be associated with a two-dimensional array of cells, conventionally arranged in a plurality of rows and columns. Row and column selection circuits conventionally enable selection of one row and one column of the two-dimensional array, so as to select one memory cell to be sensed. The two-dimensional array of memory cells, the row selection circuits and the column selection circuits are not shown in the drawings and will not be described in detail since they are considered per-se known. 
     The sensing circuit is of the so-called differential type and comprises two circuit branches  11 ,  12 . 
     Circuit branch  11 , also referred to as memory cell circuit branch, includes a plurality of parallelly connected, selectively activatable P-channel MOSFETs MP 1 , an N-channel MOSFET MN 1  and a memory cell column or bit line MBL of the two-dimensional array of memory cells. MOSFETs MP 1  have source electrodes commonly connected to a supply voltage VDD, for example the supply voltage of the memory device of 5 V or 3 V or less, and drain electrodes commonly connected to a drain electrode DN 1  of MOSFET MN 1 . MOSFET MN 1  has a source electrode SN 1  which can be electrically coupled to the memory cell bit line MBL. The column selection circuits, not shown in the drawing because per-se conventional, enable selectively coupling to circuit branch  11  one selected column of the plurality of columns of the two-dimensional array, that is, for selectively coupling the source electrode SN 1  of MOSFET MN 1  to the selected memory cell bit line MBL. 
     Circuit branch  12 , also referred to as reference circuit branch, includes a P-channel MOSFET MP 2  an N-channel MOSFET MN 2  and a reference cell bit line RBL. MOSFET MP 2  has a source electrode connected to the supply voltage VDD and a drain electrode connected to a drain electrode DN 2  of MOSFET MN 2 . MOSFET MN 2  has a source electrode SN 2  electrically coupled to the reference bit line RBL. 
     Both the memory cell bit line MBL and the reference cell bit line RBL include a plurality of memory cells MC 1 , MC 2 , . . . and RC 1 , RC 2 , . . . , respectively. In each of the bit lines MBL and RBL the memory cells are connected in parallel to each other. 
     In the shown and herein described example the memory cells and the reference cells are floating-gate MOS transistors, of the type used in EPROM, EEPROM and Flash EPROM or EEPROM memories. A floating-gate MOS transistor is a MOS transistor having an externally-accessible control gate electrode and a floating gate electrode in which charges can be trapped during a memory cell programming phase. However, this is not to intended as limitative to the present invention, since the memory cells could for example be simple MOSFETs of the type used in mask ROMs. 
     The memory cells and the reference cells have each a drain electrode connected to the respective bit line, a source electrode connected to a reference potential (ground) and the control-gate electrode connected to one of a plurality of word lines WL 1 , WL 2 , . . . of the two-dimensional array of memory cells. 
     The memory cells MC 1 , MC 2 , . . . are assumed to be multi-level memory cells, each having for example four possible threshold voltage levels and being thus capable of storing two bits of information. However, this is not to be intended as a limitation of the present invention, which is in general applicable to the sensing of memory cells having more or less than four possible threshold voltage levels. In particular, the invention can be applied also to the sensing of two-level memory cells, having two possible threshold voltage levels and being thus capable of storing one bit of information, albeit the advantages of the sensing circuit of the invention are best appreciated in connection with the sensing of memory cells having more than two possible threshold voltage levels. 
     Gate electrodes of MOSFETs MP 1  are commonly connected to the drain electrode of MOSFET MP 2 , and a gate electrode of MOSFET MP 2  is connected to the drain electrode of MOSFETs MP 1 . Because of this arrangement, MOSFETs MP 1  and MP 2  form a current mirror and bistable latch structure. 
     A bias voltage generator  13  generates a gate bias voltage V GC  for biasing gate electrodes of MOSFETs MN 1  and MN 2 . 
     The memory cell and reference circuit branches  11 ,  12  have respective output nodes N 1 , N 2  connected to respective inputs of a voltage comparator  14 . An output of the voltage comparator  14  controls a successive approximation register (“SAR”)  15 . The successive approximation register  15  generates control signals  16  controlling the selective activation of MOSFETs MP 1  so as to vary a current mirroring factor of the current mirror structure formed by MOSFETs MP 1  and MP 2 . 
     The successive approximation register  15  may for example implement a dichotomic search algorithm, as will be better explained later on in this description. 
     An output  16  of the successive approximation register  15  provides a digital code representative of the datum stored in the sensed memory cell. 
     A voltage limiter circuit, in the shown example formed by a pair of diodes D 1 , D 2  in reverse parallel connection, is connected between the output nodes N 1  and N 2  to limit a voltage swing of one of the output nodes with respect to the other. 
     Let it be assumed that the selected memory cell to be sensed, so as to read the datum stored therein, is memory cell MC 1 . The row selection circuits biases the word line WL 1  to which memory cell MC 1  belongs to a prescribed read voltage VG, for example equal to 6 V, while keeping the remaining word lines WL 2 , and so forth, of the two-dimensional array to ground. In this way, memory cell MC 1  and reference cell RC 1  are selected. 
     MOSFET MN 1  and the selected memory cell MC 1 , as well as MOSFET MN 2  and the selected reference cell RC 1  form respective cascode circuit configurations inserted in the circuit branches  11  and  12 , respectively. The cascode circuit configuration has a rather high output impedance (the impedance seen at nodes N 1  and N 2 ) and a reduced Miller effect, which enables decoupling the output nodes N 1  and N 2  from the rather high parasitic capacitances C 1  and C 2  respectively associated with the bit lines MBL and RBL. This reduces the time constant associated with the output nodes N 1  and N 2  of the two circuit branches  11  and  12 . Additionally, MOSFETs MN 1  and MN 2  bias the respective bit line MBL and RBL, and thus the drain electrode of the memory cell MC 1  to be sensed and of the reference cell RC 1 , to a value equal to 
     
       
           V   GC   −V   T −(2 I /β) ½   
       
     
     where V CG  is the value of the gate bias voltage V CG , V T  is the threshold voltage of MOSFETs MN 1  and MN 2 , I is the current flowing through MOSFETs MN 1  or MN 2  and β is a factor depending on the aspect ratio W/L (ratio of channel width W to channel length L) of the MOSFETs. The drain potential of the memory cell and of the reference cell is thus determined by the value V CG  of the gate bias voltage V CG  of MOSFETs MN 1  and MN 2  on the threshold voltage of such MOSFETs and on the current flowing therethrough. 
     In order to perform the desired function, MOSFETs MN 1  and MN 2  have to operate in saturation condition; this puts a constraint on the value of the gate bias voltage V CG . Another constraint is derived from the fact that the drain potential of the memory cell to be sensed and of the reference cell is to be kept sufficiently low not to induce spurious injection of charges into the floating gate electrodes of the memory cells and of the reference cells. 
     The reference cells RC 1 , RC 2 , and so forth, are programmed in a predetermined condition, that is the amount of charge trapped on the floating-gate electrodes thereof is known a priori. Consequently, the reference cells have a well-known threshold voltage level. The current Ir sunk by the reference cell RC 1  belonging to the selected word line WL 1  is thus known and is used as a reference current to be compared to the a priori unknown current Ic sunk by the selected memory cell MC 1  to be sensed. 
     The current Ic sunk by the selected memory cell MC 1  depends on the threshold voltage level thereof, and can thus vary from a relatively high current to a low or even zero current. 
     MOSFETs MP 1  and MP 2  behave as current-voltage converters, converting respective currents IC and IR flowing therethrough into voltage signals at nodes N 1  and N 2 , respectively. The difference between the voltage signal at nodes N 1  and N 2  depends on the current mirroring ratio of the current mirror structure, that is on the number of MOSFETs MP 1  which are activated. 
     The voltage signals at nodes N 1  and N 2  are compared by the comparator  14 . 
     In the non-limitative example of four-level memory cells, which have four possible threshold voltage levels, a memory cell biased in the prescribed sense condition sinks one of four different current values. Under the control of the successive approximation register  15  three different current mirroring factors can be programmed in the sensing circuit, by selectively activating MOSFETs MP 1 . In this way, by means of only one reference current IR, it is possible to discriminate among the four different current values that can be sunk by the memory cell. 
     As previously mentioned, the algorithm implemented by the successive approximation register can advantageously be dichotomic. This algorithm is made up of two steps. A first step provides for discriminating whether the memory cell current is one of the two higher possible current values, corresponding to the two lower threshold voltage levels, or it is one of the two lower possible current values, corresponding to the two higher threshold voltage levels. Once this has been ascertained, a second step provides for discriminating which one of the two higher or lower possible current values is the memory cell current. 
     The dichotomic algorithm has the advantage of providing in any case a result in two steps, but it is not the only possible algorithm. Another possible algorithm is for example the one providing for firstly discriminating between the two lower possible current values, then between these and the two higher possible current values, and finally discriminating between the two higher possible current values. Compared to the previous algorithm, this one has the disadvantage that the result can be provided already at the first step, if the memory cell current is the lowest possible current value, or at the second step, or at the third step. 
     It is to be noted that the variable mirroring ratio structure of FIG. 1 is only one possible implementation of sensing circuit for sensing multi-level memory cells and the present invention is not to be intended as limited to this implementation. In another possible implementation, a fixed current mirroring ratio is provided for in the two branches  11  and  12 , and a variable reference current generator is provided in the reference circuit branch  12 . The variable reference current generator, under the control of the successive approximation register, generates different reference currents to be compared to the memory cell current. 
     In order to assure that MOSFET MN 1  works properly for any current sunk by the selected memory cell to be sensed, an additional or bias current is made to flow through MOSFET MN 1 . To this purpose, a first current generator Iin 1  or first current injector injects a current Iin 1  into the drain electrode DN 1  of MOSFET MN 1 , while a second current generator Iout 1  or first current extractor extracts a current Iout 1  from the source electrode SN 1  of MOSFET MN 1 . 
     Provided that the values Iin 1 , Iout 1  of the currents injected into the drain electrode and extracted from the source electrode of MOSFET MN 1  are such that Iout 1 =Iin 1 , the bias current made to flow through MOSFET MN 1  has no effects on the value of the current IC flowing in the upper portion of circuit branch  11 , that is through MOSFETs MP 1 . In other words, the bias current flows exclusively through MOSFET MN 1  and the current IC is still equal to the current Ic sunk by the memory cell to be sensed. The bias current made to flow through MOSFET MN 1  biases MOSFET MN 1  and sets the working point thereof so as to guarantee the operation in saturation condition for any value of the current Ic sunk by the memory cell to be sensed. In particular, the bias current made to flow through MOSFET MN 1  guarantees that the latter operates in saturation condition also in the case the memory cell to be sensed sinks a low or even zero current. 
     The bias current made to flow through MOSFET MN 1  makes the latter less sensitive to the differences in the current Ic sunk by the memory cell to be sensed. This consequently reduces the variations in the drain voltage of the memory cell to be sensed and makes the biasing condition of the latter less dependent on the threshold voltage level thereof. 
     Additionally, by making the bias current to flow through MOSFET MN 1  the transconductance thereof is increased. This reduces the transient required for charging the parasitic capacitance C 1  associated with the selected bit line MBL. In fact, the transconductance of MOSFET MN 1  is proportional to the square root of the current flowing therethrough, and the time constant associated with the bit line MBL is inversely proportional to the sum of the memory cell transconductance and the transconductance of MOSFET MN 1 . 
     In the choice of the value of the bias current, the following consideration has to be made. On one hand, a high bias current value is advantageous since it causes MOSFET MN 1  to operate in strong saturation, determines a reduction in the variations in the drain voltage of the memory cell to be sensed and shortens the transient for charging the parasitic capacitance C 1  of the bit line MBL. On the other hand, a high bias current value means a high current consumption which impacts the overall consumption of the memory device, since it is to be kept in mind that in a memory device a large number of sensing circuits are normally provided for. 
     Additional considerations for the choice of the value of the bias current stem from the observation that in practice it is almost impossible to make the current Iout 1  extracted by current generator Iout 1  from the source electrode SN 1  of MOSFET MN 1  exactly equal to the current Iin 1  injected into the drain electrode DN 1  by current generator Iin 1 . A difference ΔI=Iout 1 −Iin 1  in the two currents will normally exist. Such a current difference has a direct impact on the current IC flowing through the upper portion of circuit branch  11 , whose value is no longer coincident with the value Ic of the current Ic sunk by the memory cell to be sensed and becomes IC=Ic+ΔI. 
     A first consequence of this is that, in order not to alter the matching between the two branches  11 ,  12  of the sensing circuit, an ideally equal bias current shall be made to flow through MOSFET MN 2  in the reference branch  12 . To this purpose, a third current generator or second current injector Iin 2  is provided to inject a current Iin 2  into the drain electrode DN 2  of MOSFET MN 2  and a fourth current generator or second current extractor Iout 2  is provided to extract a current Iout 2  from the source electrode SN 2  of MOSFET MN 2 . In this way, provided that Iout 2 −Iin 2 =Iout 1 −Iin 1 ≡ΔI, the value IR of the current IR flowing through the upper portion of circuit branch  12  is no longer coincident with the current Ir sunk by the selected reference cell and becomes IR=Ir+ΔI. 
     The necessity of providing a bias current also for MOSFET MN 2  has however the advantage of shortening the transients required for charging the parasitic capacitance C 2  associated with the reference bit line RBL, for the same reasons previously explained in connection with MOSFET MN 1 . 
     As a second consequence, the practically unavoidable existence of a non-zero value of ΔI suggests keeping the bias currents for MOSFETs MN 1 , MN 2  low, since the value of the current difference ΔI depends on the absolute values of Iin 1 , Iout 1 , Iin 2  and Iout 2 . 
     Other consequences of the existence of a difference ΔI in the values of the pairs of currents Iin 1 , Iout 1  and Iin 2 , Iout 2  impact the design of the current generators Iin 1 , Iout 1 , Iin 2  and Iout 2 , as will be explained below. 
     First of all, for the sake of explanation let it be assumed that Iin 1  is substantially zero and Iout 1  is much greater than the difference in the current sunk by a memory cell corresponding to two adjacent threshold voltage levels. This corresponds to having a difference ΔI&gt;0. In this condition the sensing circuit does not work correctly. Due to the presence of the current Iout 1 , a current IC=Iout 1  will flow even in the case the memory cell to be sensed has the highest threshold voltage level and does not sink any current. Such a memory cell will thus be erroneously considered as having a lower threshold voltage and the datum stored therein will not be read correctly. A difference ΔI&gt;0 can thus mask the lowest memory cell currents, that is the currents sunk by memory cells having the highest threshold voltage values. On the contrary, a difference ΔI&lt;0 does not mask the memory cell currents, the effect being at most an expansion of the current dynamic. 
     Therefore, the current generators Iin 1 , Iout 1 , Iin 2  and Iout 2  are preferably designed in such a way to assure that the differences Iout 1 −Iin 1  and Iout 2 −Iin 2  are negative. 
     Incidentally, it is worth observing that by making Iout 1 −Iin 1 , negative the working point of the memory cell to be sensed is moved away from the saturation region and more reliably kept in the ohmic region. Because of this, higher variations in the value of the read voltage VG can be withstood, since in the ohmic region the variations in the transconductance are lower. 
     Another requirement that the current generators Iin 1 , Iout 1 , Iin 2  and Iout 2  will satisfy is the stability in temperature of the current differences Iout 1 −Iin 1  and Iout 2 −Iin 2 . 
     In the design of the current generators, particularly the current extractors Iout 1  and Iout 2  which extract the currents Iout 1  and Iout 2  from the source electrodes SN 1  and SN 2  of MOSFETs MN 1  and MN 2  care must be put so that current generators Iout 1  and Iout 2  work correctly also in the case the voltage drop across them, that is the drain voltage of the memory cell to be sensed and the selected reference cell, falls to low values. 
     Additionally, the output resistance of the current generator Iout 1  shall be sufficiently high to prevent an undesirable increase of the current Iout 1  in consequence of a rise of the memory cell drain voltage during the programming thereof. In the programming phase the drain electrodes of the memory cells are in fact brought to a relatively high voltage value, for example 5 V or more, to induce injection of charges into the floating gate electrodes thereof. 
     The results of experimental trials conducted by the Applicant are reported hereinbelow. Assuming that the bit line MBL has a parasitic capacitance C 1  of approximately 1 pF, and that a memory cell programmed in the lowest threshold voltage level sinks approximately 50 μA, if no bias current is provided for, the transconductance of MOSFET MN 1  is approximately equal to 400 μA/V and the time constant associated with the source electrode SN 1  of MOSFET MN 1  is approximately 2 ns. In the same conditions, assuming that a memory cell programmed in the highest threshold voltage level (fully programmed memory cell) sinks no current at all, MOSFET MN 1  is off and its transconductance is therefore approximately zero; the time constant associated with the source electrode SN 1  becomes approximately equal to 100 ns, which makes the time required for sensing the memory cell unacceptably long. By providing a bias current of approximately 5 μA to flow through MOSFET MN 1 , also in the case of a fully programmed memory cell MOSFET MN 1  is kept turned on and the transconductance thereof is approximately equal to 125 μA/V, so that the time constant associated with the source electrode thereof is less than approximately 8 ns. 
     FIGS. 2 and 3 show corresponding portions of sensing circuits according to further embodiments of the present invention. The differences of these embodiments with respect to the one previously described reside only in the circuit for controlling MOSFETs MN 1  and MN 2 . For symmetry of the structure, only a portion of the memory cell circuit branch  11  is depicted in FIGS. 2 and 3. 
     More specifically, in FIG. 2 MOSFET MN 1  is controlled by a feedback network including an inverter INV 1 , for example a CMOS inverter, with an input connected to the source electrode SN 1  of MOSFET MN 1  and an output connected to the gate electrode of MOSFET MN 1 . The inverter is designed to work in the linear region of its input-output characteristic, and the inverter threshold voltage determines the value of the voltage at the source electrode SN 1  of MOSFET MN 1 , that is the drain potential of the memory cell to be sensed. 
     In FIG. 3 MOSFET MN 1  is still controlled by a feedback network, as in FIG.  2 . However, in this case the feedback network includes an operational amplifier OP 1  with an inverting input connected to the source electrode SN 1  of MOSFET MN 1 , and a non-inverting input supplied with a reference voltage V R  generated by a reference voltage generator  30 . An output of the operational amplifier  30  is connected to the gate electrode of MOSFET MN 1 . The value of the reference voltage V R  determines the value of the voltage at the source electrode SN 1  of MOSFET MN 1 G, that is the value of the drain potential of the memory cell to be sensed. 
     FIG. 4 shows a circuit diagram of a preferred embodiment of current generators In 1 , Iout 1 , Iin 2 , Iout 2 . The two current generators Iout 1  and Iout 2  or current extractors are substantially identical in structure and each comprise a first circuit branch  41 ,  42  and a second circuit branch  43 ,  44  connected to form a current mirror. The first circuit branch  41 ,  42  includes two serially-connected N-channel MOSFETs M 5  and M 4 , respectively M 12  and M 11 . MOSFET M 4 , respectively M 11 , has a source electrode connected to ground and a gate electrode connected to a drain electrode of MOSFET M 5 , respectively M 12 . The second circuit branch  43 ,  44  includes one N-channel MOSFET M 3 , respectively M 10 , with a source electrode connected to ground and a gate electrode connected to the gate electrode of MOSFET M 4 , respectively M 11 , in the first branch. The first branch  41 ,  42  is run through by a reference current Iref 1 , respectively Iref 2 , which in the case of current generator Iout 1  is generated by a reference current generator Iref while in the case of current generator Iout 2  is derived from current Iref 1  by current mirroring, as will be explained later on. An operational amplifier  45 ,  46  has an inverting input connected to a drain electrode of MOSFET M 4 , respectively M 11 , in the first branch  41 ,  42 , and a non-inverting input connected to a drain electrode of MOSFET M 3 , respectively M 10 , in the second branch  43 ,  44 . An output of the operational amplifier  45 ,  46  controls a gate electrode of MOSFET M 5 , respectively M 12 . The currents Iout 1  and Iout 2  are sunk by MOSFETs M 3  and M 10 , respectively, which have the drain electrodes thereof connected to the source electrode SN 1  and SN 2  of MOSFETs MN 1  and MN 2  respectively. 
     The current Iref 1  is mirrored into a circuit branch  47 , comprising two serially-connected N-channel MOSFETs M 2 , M 1 . MOSFET M 1  has a source electrode connected to ground and a gate electrode connected to the gate electrode of MOSFET M 4 . MOSFET M 2  has a gate electrode connected to the gate electrode of MOSFET M 5 , that is controlled by the output of the operational amplifier  45 . Circuit branch  47  also comprises, in series to MOSFETs M 2  and M 1 , two serially-connected P-channel MOSFETs M 9  and M 6 . MOSFET M 9  has a source electrode connected to the supply voltage VDD and a gate electrode connected to a drain electrode of MOSFET M 6 . A gate electrode of MOSFET M 6  is connected to a reference voltage generator Vref generating a bias voltage. 
     The current flowing through branch  47  is in turn mirrored into circuit branch  42  by means of two serially-connected P-channel MOSFETs M 16  and M 15  with respective gate electrodes connected to the gate electrodes of MOSFETs M 9  AND M 6 , respectively. 
     Current generators In 1  and In 2  are formed by two further circuit branches  48 ,  49 , each comprising two serially-connected P-channel MOSFETs M 8  and M 7 , respectively M 14  and M 13 , with respective gate electrodes connected to the gate electrodes of MOSFETs M 9  and M 6 , respectively. The currents Iin 1  and Iin 2  are respectively delivered by MOSFETs M 7  and M 13 , whose drain electrodes are connected to the drain electrodes of MOSFETs MN 1  and MN 2  respectively. 
     Considering the current generator Iout 1 , the output voltage of the operational amplifier  45  is given by V T (M 5 )+V ovd (M 5 )+V DS (M 4 ), where V T (M 5 ) is the threshold voltage of MOSFET M 5 , V ovd (M 5 ) is the voltage overdrive of MOSFET M 5  and V DS (M 4 ) is the voltage across MOSFET M 4 . The difference between the reference current value Iref 1  and the current Iout 1  sunk by MOSFET M 3  is thus found to be:            Iref                 1       Iout                 1       =       (     1   +     λ   ·       V   DS          (     M                 4     )           )       (     1   +     λ   ·     (           A   +   1     A            V   DS          (     M                 4     )         +           V   T          (     M                 5     )       +       V   ovd          (   M5   )         A       )         )                              
     where A is the gain of the operational amplifier  45  and λ is the channel-length modulation parameter expressing the Early effect in MOSFET M 4 . This structure has the advantage of being capable of working also for low or almost zero voltages across MOSFET M 4 , which coincides with the drain voltage of the memory cell to be sensed. The difference ΔI between current Iout 1  and current Iin 1 , the latter coincident with current Iref 1  due to the current mirroring, is only affected by the mismatches between the circuit components and the offset of the operational amplifier  45 . Additionally, by means of a small-signal analysis it is possible to realize that the output resistance of the current generator Iout 1  is approximately equal to the operational amplifier gain A times the output resistance of MOSFET M 3  which means, due to the high value of A, a very high value. 
     FIG. 5 is a diagram reporting the result of simulations conducted by the Applicant on the circuit structure of FIG.  4 . The abscissa of the diagram is the value of the drain potential of the memory cell MC 1 , in volts. It is possible to see that the currents Iin 1  and Iout 1 , respectively injected into the drain electrode DN 1  and extracted from the source electrode SN 1  of MOSFET MN 1 , are substantially equal to each other and to approximately 5 μA even for very low values of the memory cell drain potential. 
     Although the present invention has been disclosed and described by way of some embodiments, it is apparent to those skilled in the art that several modifications to the described embodiments, as well as other embodiments of the present invention are possible without departing from the spirit or essential features thereof.