Abstract:
A DC-DC converter, which controls the output voltage supplied to a load at a desired magnitude by performing on/off control of the input voltage using a switch, includes: an error amplifier for outputting the difference voltage between the output voltage and a preset reference voltage; and a plurality of phase compensation circuits for compensating the phase of the output voltage fed back to the error amplifier with different characteristics, whereby the DC-DC converter is configured such that changes in either the input voltage or in the load current flowing into the load are detected, and switching between the plurality of phase compensation circuits is performed. The frequency characteristic of each of the phase compensation circuits is determined for each of a plurality of demarcated fluctuation ranges of the input voltage or the load current.

Description:
BACKGROUND OF THE INVENTION 
       [0001]    The invention relates to a DC-DC converter which changes a load current supplied to a load circuit by performing on/off control of a DC input voltage using a switching device to control a DC output voltage at a desired magnitude, and in particular relates to a DC-DC converter suitable for use as a DC power supply apparatus for portable electronic equipment using a rechargeable battery as a driving power supply. 
         [0002]    In portable telephones and other electronic equipment of the prior art, rechargeable lithium batteries and similar rechargeable batteries are installed as power sources. In general, the output voltage of a battery declines depending on circumstances of equipment usage, battery discharge, etc., and so a DC-DC converter is provided to convert direct current into direct current at an arbitrary voltage (DC-DC conversion) in order to convert the battery voltage into a constant voltage for output. However, even when using the same DC-DC converter, depending on the electronic equipment, the voltage input to the DC-DC converter is different depending on the lithium battery specifications (for example, whether a number of lithium battery cells are connected in series). On the other hand, in electronic equipment that operates on a battery, mode-switching is performed into a mode in which a charged battery is maintained for a long period of time, and current consumption is reduced in order to lengthen the operation time of the electronic equipment. To this end, not only does the voltage difference between the input voltage and the output voltage change according to the battery specifications of the applied electronic equipment and the circumstances of use of the electronic equipment, but the load current itself also changes according to the state of use of the equipment. 
         [0003]    In the past, DC-DC converters have been used in various electrical and electronic equipment, and DC-DC converters have been proposed according to the purpose of use, such as step-up DC-DC converters and step-down DC-DC converters having feedback loops employing voltage mode control or current mode control, and similar. 
         [0004]      FIG. 8  is a circuit diagram showing a step-down type DC-DC converter of the prior art. This DC-DC converter comprises, within a control IC  1 , a reference voltage generator  11 , internal power supply  12 , an error amplifier  13 , preferably comprising an operation amplifier, a sawtooth wave generator  14 , a PWM comparator  15 , and an output MOS driver  16 . A switch Q 1 , comprising a MOS transistor, is turned on and off by digital signals from the output MOS driver  16 . An input voltage Vcc, with an input range of for example 2.5 V to 10 V, is supplied to the switch Q 1 , and when the switch Q 1  performs switching operation, an output voltage Vo (for example, 2 V) lower than the input voltage Vcc is supplied to the load Ro by means of a commutation diode (flywheel diode) D, inductor L, and output smooth capacitor Cout. 
         [0005]    At this time, in order to stabilize the output voltage Vo supplied to the load Ro through switching operation, a feedback voltage Vfb, which is normally obtained by voltage-dividing using sensing resistors Ra, Rb or similar, is supplied as negative feedback to the control IC  1 . In the control IC  1 , the error between the feedback voltage Vfb and the reference voltage Vref is amplified by the error amplifier  13 , and is compared with the sawtooth wave signal voltage Vtr from the sawtooth wave generator  14  by the PWM comparator  15 , the result is converted into a PWM (Pulse Width Modulation) signal which controls the time ratio (duty), and the signal from the output MOS driver  16  performs on/off control of the switch Q 1 . 
         [0006]    As a result, the error voltage Verr output from the error amplifier  13  is controlled so as to increase when the feedback voltage Vfb is smaller than the reference voltage Vref and, conversely, to decrease when Vfb is larger than Vref, and by controlling the duty ratio of the switch Q 1 , the output voltage Vo is controlled. 
         [0007]    The characteristic of such a DC-DC converter with voltage-mode control is such that oscillation readily occurs due to second-order lag of the main circuit comprising the inductor L and output smoothing capacitor Cout, and so phase compensation is indispensable in order to effect stable operation of the negative-feedback circuit. A general explanation relating to phase compensation to prevent oscillation in error amplifiers is given in  OS - CON Conductive Polymer - Aluminum Solid Electrolytic Capacitors, Organic Semiconductor - Aluminum Solid Electrolytic Capacitors , Technical Book Ver. 14 (online), October 2006, SANYO Electronic Device Company (search date Aug. 31, 2007), &lt;URL: HYPERLINK “http://edc.sanyo.com/pdf/oscon/OS” http://edc.sanyo.com/pdf/oscon/OS_Jpdf&gt;. 
         [0008]    If the capacitance value of the output capacitor is increased, the DC-DC converter is easily stabilized, but cost becomes a problem. A method may also be used of providing a resistance circuit in series with the output capacitor to advance the phase, but this is not desirable due to the fact that the power conversion efficiency of the DC-DC converter is lowered. As indicated in the above-referenced document currently the most widespread phase compensation employs, as a phase lag compensation circuit for the error amplifier  13 , the series circuit of a resistor R 1  and capacitor C 1  connected between an inverting input terminal to which the feedback voltage Vfb is input and an output terminal, and, as a phase lead compensation circuit, the series circuit of a resistor R 2  and capacitor C 2  connected in parallel with the sensing resistor Ra. 
         [0009]    In Japanese Patent Laid-open No. H05-304771 (paragraphs [0009] to [0017],  FIG. 1 ), a circuit is disclosed in which, in contrast with continuous mode in which current always flows in the choke coil, because in discontinuous modes in which current flows intermittently in the choke coil below a critical point the DC-DC converter transfer function changes, a configuration is employed in which a feedback phase compensation circuit is selectively switched according to changes in the transfer function to suppress phase rotation. 
         [0010]    Further, in Japanese Patent Laid-open No. 2006-304552 (paragraphs [0017] to [0066],  FIG. 1 ), a circuit is disclosed in which a voltage follower and gain-adjusting resistor are provided between a voltage-dividing resistor and the error amplifier, and the resistance value of the gain-adjusting resistor is adjusted so as to be inversely proportional to the output voltage of the switching regulator, to hold constant the gain of the error amplifier stage. 
         [0011]    Because the phase characteristic of a DC-DC converter varies greatly with the input voltage Vcc, output voltage Vo, load current Io, and other parameters, each of the constants of the phase compensation circuit must be set optimally taking these conditions into consideration. Further, for some applications of the DC-DC converter, it is also necessary to secure adequate response characteristics upon startup, rapid changes in the input voltage Vcc and load current Io, changes in the output voltage Vo, and similar. 
         [0012]    However, in a DC power supply device for portable electronic equipment in which a rechargeable lithium battery or similar is used, the fluctuation ranges of the input voltage Vcc and load current Io of the DC-DC converter are set extremely broadly, and there are cases in which a simple combination of sensing compensation and error amplification compensation alone is insufficient to obtain adequate stability and response characteristics. 
       SUMMARY OF THE INVENTION 
       [0013]    The invention was devised in light of the above-described problems, and provides a DC-DC converter with a broad usage range as a DC power supply device, and in particular one which is capable of optimal phase compensation with a simple configuration, even when there are changes in the input voltage and load current. 
         [0014]    More specifically, in order to resolve the above problems, a DC-DC converter is provided, which controls the DC output voltage supplied to a load circuit at a desired magnitude by performing on/off control of the DC input voltage using a switching device, including an error detection device, which outputs the difference voltage between the DC output voltage and a preset reference voltage; a phase compensation device, including a plurality of phase compensation circuits which use different characteristics to perform compensation of the phase of the DC output voltage fed back to the error detection device; and a switching control device, which detects changes in either the DC input voltage or in the load current flowing into the load circuit, and switches between the plurality of phase compensation circuits constituting the phase compensation device; and is characterized in that each of frequency characteristics of the phase compensation circuits is determined for each of a plurality of demarcated fluctuation ranges of the DC input voltage or the load current flowing into the load circuit. 
         [0015]    In this DC-DC converter, switching control is performed such that the phase compensation circuit is switched to one phase compensation circuit when the input voltage or load current falls below a reference value, and is switched to another phase compensation circuit when the value rises above the reference value. 
         [0016]    Accordingly, by means of this invention, phase compensation circuit constants can easily be set, and moreover the output voltage can be set with stability over all regions even when the input voltage range and load current range are broad. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0017]    The invention will now be described with reference to certain preferred embodiments thereof and the accompanying drawings, wherein: 
           [0018]      FIG. 1  is a circuit diagram showing a DC-DC converter in accordance to a first embodiment of the invention; 
           [0019]      FIG. 2  is a graph showing the single-loop gain frequency characteristic (amplitude frequency characteristic) of the DC-DC converter shown in  FIG. 1 ; 
           [0020]      FIG. 3  is a circuit diagram showing the DC-DC converter according to a second embodiment of the invention; 
           [0021]      FIG. 4  shows the (first) results to determine the single-loop frequency characteristic of a DC-DC converter; 
           [0022]      FIG. 5  shows the (second) results to determine the single-loop frequency characteristic of a DC-DC converter; 
           [0023]      FIG. 6  shows the (third) results to determine the single-loop frequency characteristic of a DC-DC converter; 
           [0024]      FIG. 7  shows the (fourth) results to determine the single-loop frequency characteristic of a DC-DC converter; and 
           [0025]      FIG. 8  is a circuit diagram showing a step-down DC-DC converter of the prior art. 
       
    
    
     DESCRIPTION OF THE PREFERRED EMBODIMENTS 
       [0026]    Below, preferred embodiments of the invention are explained referring to the drawings.  FIG. 1  is a circuit diagram showing the DC-DC converter according to a first embodiment. Here, circuit elements corresponding to those of the circuit of the prior art in  FIG. 8  are assigned the same symbols, and explanations thereof are omitted. 
         [0027]    In this aspect, compared with the DC-DC converter of  FIG. 8 , a comparator  2  which outputs a switching control signal SWcnt, series resistors Rc, Rd, Re which set the reference voltages V 1 , V 2  of this comparator  2 , series resistors R 10 , R 20  to voltage-divide the input voltage Vcc and generate a voltage signal Vcd which is supplied to the non-inverting input terminal of the comparator  2 , a switch SW, and a capacitor C 3  which is selected by the switch SW, are added. The comparator  2 , series resistors Rc, Rd, Re, series resistors R 10 , R 20 , and switch SW forming switching control device. One end of the series resistors Rc, Rd, Re is connected to the constant voltage Vref or Vreg, and the other end is grounded. The potential V 1  at the connection point of resistors Rc and Rd and the potential V 2  at the connection point of resistors Rd and Re are input to the two inverting input terminals of the comparator  2  as reference voltages. The comparator  2  uses the two reference voltages V 1  and V 2  to operate as a hysteresis comparator. That is, the comparator  2  has a hysteresis characteristic such that, when the switching control signal SWcnt changes from H (high) to L (low), V 2  becomes the reference voltage, and when the signal changes from L to H, V 1  becomes the reference voltage. When the comparator  2  need not operate as a hysteresis comparator, the resistor Re may be omitted, so that only the reference voltage V 1  is input to the comparator  2 . The switch SW is controlled by switching control signals SWcnt of the comparator  2 . 
         [0028]    Here, the range of the input voltage Vcc for the DC-DC converter is 2.5 to 10 V, and moreover the load current Io flowing in the load Ro is assumed to vary in the range 0 to 1 A. A case is assumed in which, as explained above, in the prior art stable operation of the control IC  1  cannot be obtained over the entire range of the input voltage Vcc and load current Io. 
         [0029]    Hence a case is here studied in which the input voltage Vcc initially changes in a low voltage range (2.5 to 6 V), and the resistance values and capacitance values (that is, capacitances C 1  and C 2  and resistances R 1  and R 2 ) for the optimal phase compensation circuit when the load current Io is from 0 to 1 A are determined. Then, for a high voltage range (4 to 10 V), upon similarly varying the load current Io between 0 and 1 A, with the capacitance C 1  and the resistances R 1  and R 2  fixed, the capacitance value of capacitor C 2  is studied. By this means, the value of a new capacitor C 3  constituting the optimal phase compensation circuit can be determined. 
         [0030]    The specific capacitance values and similar in a phase compensation circuit can be determined by analysis using well-known state averaging equations of the prior art, such as for example directly analyzing state averaging equations, or, when this is difficult, performing analyses using simulations (for information on state averaging methods, see for example Kousuke Harada et al,  Fundamentals of Switching Converters  (Corona Publishing Co., Ltd., 1992)). Or, experiments may be performed in advance to determine the resistance values of resistors, capacitance values of capacitors, and similar which are to be modified. 
         [0031]    The resistance values of the series resistors Rc, Rd, Re are set so as to supply reference voltages V 1  and V 2  equal to the voltage values of voltage signals Vcd corresponding to the voltage values of the input voltage Vcc at which the high-voltage range and low-voltage range overlap (for example, 5.5 V and 4.5 V) (when there is no hysteresis, the values are set so as to supply V 1 , equal to the voltage signal Vcd corresponding to Vcc=5.0 V). A switching control signal SWcnt for the phase compensation circuit is supplied from the comparator  2  so as to switch the switch SW to the side of capacitor C 2  when the input voltage Vcc falls to 4.5 V or lower, and so as to switch the switch SW to the side of capacitor C 3  when the input voltage Vcc rises to 5.5 V or higher. 
         [0032]    Fluctuations in the input voltage range will be explained with reference to  FIG. 2 .  FIG. 2  is a graph showing the single-loop (open-loop) gain frequency characteristic (amplitude frequency characteristic) for the DC-DC converter shown in  FIG. 1 . However, here there is no switching between phase compensation circuits of the phase compensation device. 
         [0033]    In  FIG. 2 , Vcc 1 , indicated by a solid line, and Vcc 2 , indicated by a broken line, are two input voltages to the DC-DC converter; a case is shown in which Vcc 1 &gt;Vcc 2 . As shown in  FIG. 2 , the entire (single-loop) gain of the DC-DC converter for input voltage Vcc 2  is low compared with that for the input voltage Vcc 1 . That is, if no measures are taken with respect to changes in input voltage, as the input voltage declines the band (cutoff frequency) of the DC-DC converter falls. Also, the single-loop phase characteristic (change in phase lag with frequency) of the DC-DC converter does not change with the input voltage, so that the phase margin also worsens. As a result, as the input voltage falls, the startup time during transient response of the DC-DC converter is lengthened, so that the response of the DC-DC converter is degraded, and in addition there is the problem that stability worsens. 
         [0034]    Hence as explained above, by setting phase compensation circuit switching conditions in the DC-DC converter, so that the phase compensation circuit frequency characteristic is switched with an input voltage Vcc of 5 V as the borderline, then adequate response of the DC power supply apparatus can be secured over a broad range of fluctuation of the input voltage Vcc, and moreover operation can be made stable. 
         [0035]    Further, the phase compensation circuit switched device is not limited to a capacitor C 2  alone; the resistor R 2  may be switched to different values, or, the phase lead compensation circuit on the side of the sensing resistors Ra, Rb may be left unmodified, while the magnitudes of the capacitor C 1  and resistor R 1  forming the phase lag compensation circuit for the error amplifier  13  may be modified. Or, a number of, or all of, the plurality of devices of the phase compensation circuit (capacitors C 1  and C 2 , resistors R 1  and R 2 ) may be switched at once. In essence, the values of switched devices may be selected such that the DC-DC converter frequency characteristic is stable over the entire range of fluctuation of the input voltage Vcc, which varies greatly. 
         [0036]      FIG. 3  is a circuit diagram showing the DC-DC converter according to a second embodiment of the invention. Here, a difference with the circuit of  FIG. 1  is that a load current detection resistor Rs, one end of which is grounded, is added, connected in series to the load Ro, and the voltage thereon Vs is input to the non-inverting input terminal of the comparator  3 . That is, whereas the switching control device of  FIG. 1  is configured to perform switching according to the magnitude of the input voltage Vcc to the comparator  2 , as the switching control device of  FIG. 3 , the comparator  3  performs switching according to the magnitude of the load current Io. The comparator  3 , series resistors Rf, Rg, Rh, and reference voltages V 3  and V 4  are equivalent to the comparator  2 , series resistors Rc, Rd, Re, and reference voltages V 1  and V 2  of  FIG. 1 , respectively, and perform the same respective functions. Also, explanations of cases in which hysteresis is and is not present are also similar, and so are omitted. 
         [0037]    Here also, the range of the input voltage Vcc to the DC-DC converter is from 2.5 to 10 V, and moreover the load current Io flowing in the load Ro varies in the range 0 to 1 A. As explained above, a case has been assumed in which, in the prior art, stable operation of the control IC  1  cannot be obtained over the entire ranges of the input voltage Vcc and load current Io. 
         [0038]    Here, a case is studied in which the load current Io is initially varied in a low-current range (0 to 0.6 A), and the resistance values and capacitance values (that is, the values of the capacitances C 1 , C 2  and resistances R 1 , R 2 ) of the optimal phase compensation circuit when the input voltage Vcc is varied between 2.5 and 10 V are determined. Then, in a high current range (0.4 to 1 A), the input voltage Vcc is similarly varied between 2.5 and 10 V, and with the capacitance C 1  and the resistances R 1  and R 2  fixed, the capacitance value of the capacitor C 2  is studied. By this means, the value of the new capacitor C 4  used to form the optimal phase compensation circuit can be determined. 
         [0039]    The resistance values of the series resistors Rc, Rd, Re are set so as to supply the reference voltages V 3  and V 4 , equal to the voltage Vs from the load current detection resistor Rs equivalent to the current values at which the high-current range and the low-current range of the load current Io overlap (for example, 0.55 A and 0.45 A). Here, the comparator  3  supplies a phase compensation circuit switching control signal SWcnt such that, when the voltage Vs supplied to the non-inverting input terminal of the comparator  3  is equal to or less than the reference voltage V 4 , the switch SW is switched to the side of the capacitor C 2 , and when the voltage Vs is equal to or greater than the reference voltage V 3 , the switch SW is switched to the side of the capacitor C 4 . 
         [0040]    Fluctuation of the output current range will now be explained. If the magnitude of the load current Io is replaced with the magnitude of the impedance of the load Ro, then if the impedance of the load Ro changes, the transfer function of the entire system of the DC-DC converter including the load Ro also changes, and of course the response changes. In qualitative terms, in the case of a heavy load with a large load current Io (that is, a load Ro with low impedance), phase compensation is adjusted so as to speed system response to enable supply of a large current. However, if this phase compensation circuit is left unmodified, then in the case of a light load with a small current (that is, a load Ro with high impedance), instability tends to occur. Hence it is necessary to modify the frequency characteristic of the phase compensation circuit according to the magnitude of the load current Io. 
         [0041]    Hence as explained above, by setting the phase compensation circuit switching conditions in the DC-DC converter, if the phase compensation circuit frequency characteristic is switched with a load current Io of 0.5 A as the borderline, then stable operation of the DC power supply apparatus can be attained within a broad range of fluctuation of the load current Io. Here, similarly to the case of Aspect  1 , the comparator  3  can be given a prescribed hysteresis. 
         [0042]    Next, the gain of a step-down type DC-DC converter, as well as a method of adjustment of the phase margin in the phase lag frequency characteristic, are explained based on simulation results for the frequency characteristic using the above-described state averaging method. 
         [0043]      FIG. 4  to  FIG. 7  show the results (first to fourth) of determination of the single-loop frequency characteristic of a DC-DC converter by simulations, all based on a state averaging method, indicating the phase margin when switching the capacitor of the phase lead compensation circuit between various values for an input voltage Vcc. In all of the figures, the horizontal axis indicates the frequency on a log scale, and the gain (dB) characteristic corresponding to the right-side vertical axis is indicated by a broken line, while the phase lag (deg) characteristic corresponding to the left-side vertical axis is indicated by a solid line. With the inductance L=10 μH, output smoothing capacitance Cout=4.7 μF, output voltage Vo=3 V, voltage-dividing resistances Ra=20 kΩ and Rb=10 kΩ, load resistance Ro=3Ω, and with the load current Io constant (=1 A), and under common conditions in which the phase lead compensation circuit resistance R 1 =6.2 kΩ and capacitance C 1 =470 pF and the phase lead compensation circuit resistance R 2 =1 kΩ, the respective frequency characteristics are calculated. In  FIG. 4 , the phase margin is shown when the input voltage Vcc is 5 V and the capacitance C 2  is 1 nF. Under these conditions, the phase margin of the step-down DC-DC converter is 35 degrees. However, as shown in  FIG. 5 , when the input voltage Vcc rises to 24 V, if the capacitance C 2  is left at 1 nF the phase margin shrinks to 19 degrees. 
         [0044]    Here, the phase margin is the margin of the phase lag from a phase of 180 degrees when the amplification factor is 1 (gain of 0 dB); normally, a phase margin of approximately 30 to 40 degrees is desirable. Hence the step-down DC-DC converter is in a state in which oscillation occurs readily. 
         [0045]      FIG. 5  and  FIG. 6  show frequency characteristics when the capacitance C 2  (=1 nF) of the phase lead compensation circuit is switched by the switch SW. Here, by switching to the capacitance C 3  (=150 pF), when the input voltage Vcc=5 V in  FIG. 5 , the phase margin is 24 degrees, which is smaller than that in  FIG. 4  with C 2 =1 nF, but when the input voltage Vcc rises to 24 V as in  FIG. 6 , the phase margin can be adjusted to 45 degrees. 
         [0046]    That is, when the input voltage Vcc is 5 V the capacitance is switched to C 2 =1 nF, and when the input voltage Vcc is 24 V the capacitance is switched to C 3 =150 pF. 
         [0047]    In both of the embodiments described above, examples of step-down DC-DC i  converters were explained; however, this invention can also be similarly applied to step-up type devices, or to polarity-inverting DC-DC converters, with prominent advantageous results obtained. Of course, in addition to voltage mode control, this invention can also be applied to DC-DC converters employing current mode control, in which inductor current feedback is employed. 
         [0048]    The invention has been described with reference to certain preferred embodiments thereof. It will be understood, however, that modifications and variations are possible within the scope of the appended claims. 
         [0049]    This application is based on, and claims priority to, Japanese Patent Application No: 2007-269694, filed on Oct. 17, 2007. The disclosure of the priority application, in its entirety, including the drawings, claims, and the specification thereof, is incorporated herein by reference.