Abstract:
The present invention provides a system for providing a bi-directional vector modulator in which the signal can travel in either direction while the phase shift remains constant. In addition to the functionality of being a phase shifter, the disclosed invention acts as a vector modulator by having the ability to continuously control both the amplitude and phase of a signal.

Description:
BACKGROUND OF THE INVENTION 
   a. Field of the Invention 
   The present invention relates generally to vector modulators and more particularly relates to vector modulators capable of varying the magnitude and phase of a signal that may be passing in either direction through the apparatus. 
   b. Description of the Background 
   Various methods of modulating RF signals have been developed. In particular, vector modulation in which the resultant signal is the vector sum of two amplitude modulated signals has demonstrated many advantages over other methods. These individual signals that combine to form a vector sum are phase shifted with respect to one another by a predetermined amount and may consist of a carrier signal that is modulated by a data input signal. In a typical phased array antenna system, vector modulators are employed to vary both the amplitude and phase of a signal, as opposed to merely varying the phase with a phase shifter. This type of system can reduce the amplitude of the side lobes (amplitude tapering) thereby optimizing antenna pattern. These types of antenna systems typically include a plurality of distinct antenna elements that are individually directed to transmit and/or receive a signal in a particular orientation. 
   Typical vector modulators, such as those utilized in conventional phased array antenna systems, characteristically employ amplifiers to adjust the signal gain and produce variability in the signal amplitude. Variable phase shifts and gains are obtained by adjusting the relative amplitudes of the vectors. The major drawback realized in utilizing amplifiers is that these devices are unidirectional. That is, they produce a different amplitude response and phase shift depending upon which direction the signal is being sent through the device and restrict the versatility of the circuit with respect to achieving continuous phase and amplitude states. 
   In the case of a phased array antenna system, the signal passes through the device in one direction for transmitting, and the opposite direction for receiving. Previous attempts using active phase shifters that employ a vector sum method have been subject to large power losses due to the architecture schemes of the phase shifters and inefficiencies in utilizing amplifiers. 
   Therefore, there is a need for a vector modulator capable of independently varying the magnitude and phase of a signal, and to accomplish this in either direction through the apparatus and allow amplitude tapering of the antenna. 
   SUMMARY OF THE INVENTION 
   The present invention overcomes the disadvantages and limitations of the prior art by providing a bi-directional vector modulator in which the signal can travel in either direction while the phase shift remains constant. In addition to the functionality of being a phase shifter, the disclosed invention acts as a vector modulator by having the ability to continuously control both the amplitude and phase of a signal. 
   The present invention may therefore comprise a system for bi-directionally modulating a signal and that produces a phase shift that is the same for both transmission and reception comprising: a first power divider that splits an input signal into a first output signal and a second output signal; the first output signals and the second output signals having the same phase with respect to one another; a first variable attenuator connected to the first power divider that receives the first output signal of the first power divider to generate a first attenuated signal; a second variable attenuator connecting the first power divider that receives the second output signal of the first power divider to generate a second attenuated signal, a branch line coupler connected to the first variable attenuator and the second variable attenuator that receives the first attenuated signal and the second attenuated signal; the branch line coupler producing a first coupled output and a second coupled output that are in quadrature; a phase shifter connected to the branch line coupler that receives a signal from the second output of the branch line coupler producing a shifted output signal that is phase shifted 180 degrees; a third variable attenuator connected to the branch line coupler that receives the first coupled output from the branch line coupler to generate a third attenuated signal; a fourth variable attenuator connected to the phase shifter that receives the shifted output signal from the phase shifter to generate a fourth attenuated signal; and a second power divider connected to the third variable attenuator and the fourth variable attenuator, that receives and combines the third attenuated signal and the fourth attenuated signal, that combines the third attenuated signal and the fourth attenuated signal having the same phase with respect to one another, and generates a final output signal. 
   The present invention may also comprise a method for bi-directionally modulating a signal and producing a phase shift that is the same for both transmission and reception comprising: splitting an input signal into a first output signal and a second output signal with a first power divider, the first output signal and the second output signal having the same phase with respect to one another; receiving the first output signal of the first power divider with a first variable attenuator connected to the first power divider; generating a first attenuated signal with the first variable attenuator, receiving the second output signal of the second power divider with a second variable attenuator connected to the second power divider; generating a second attenuated signal with the second variable attenuator, receiving the first attenuated signal and the second attenuated signal with a branch line coupler connected to the first variable attenuator and the second variable attenuator; producing in quadrature, a first coupled output and a second coupled output with the branch line coupler; producing a 180 degree phase shifted output signal with a phase shifter connected to the branch line coupler that receives a signal from the second output of the branch line coupler; receiving the first coupled output from the branch line coupler with a third variable attenuator connected to the branch line coupler, generating a third attenuated signal with the third variable attenuator, receiving the shifted output signal from the phase shifter with a fourth variable attenuator connected to the phase shifter; generating a fourth attenuated signal with the fourth variable attenuator; receiving and combining the third attenuated signal and the fourth attenuated signal having the same phase with respect to one another with a second power divider connected to the third variable attenuator and the fourth variable attenuator; and generating a final output signal with the second power divider. 
   The present invention may additionally comprise a phased array antenna system with N radiating/receiving elements comprising: a splitter/combiner that receives and divides an outbound RF signal into N split output signals when the antenna system is in transmit mode and that receives and combines the N inbound modulated output signals and produces a single combined inbound RF signal when the antenna system is in receive mode, a parallel circuit of N bi-directional vector modulators that receives the split output signals and produces N outbound modulated output signals when the antenna system is in transmit mode, and that receives and modulates the N inbound emitter output signals and produces the N inbound modulated output signals when the antenna system is in receive mode, a controller that controls the amplitude and phase shift introduced to the outbound modulated output signals by each of the N bi-directional vector modulators when the antenna system is in transmit mode, and that controls the amplitude and phase shift introduced to the inbound modulated output signals by each of the N bi-directional vector modulators when the antenna system is in receive mode, and an array of N radiating/receiving elements that receives the N outbound modulated output signals and produces N outbound RF waves when the antenna system is in transmit mode, and that receives N inbound RF waves and produces the N inbound emitter output signals when the antenna system is in receive mode. 
   The present invention may also comprise a method of transmitting RF signals with a phased array antenna system with N radiating/receiving elements comprising: splitting an outbound RF signal into N split output signals with a splitter/combiner; receiving the N split output signals with a parallel circuit of N bi-directional vector modulators; controlling the amplitude and phase shift introduced to the outbound modulated output signals by each of the N bi-directional vector modulators with a controller; producing N outbound modulated output signals with the N bi-directional vector modulators; receiving the N outbound modulated output signals with an array of N radiating/receiving elements; producing N outbound RF waves with the array of N radiating/receiving elements; receiving N inbound RF waves with the array of N radiating/receiving elements; controlling the amplitude and phase shift introduced to the inbound modulated output signals by each of the N bi-directional vector modulators with the controller; producing N inbound modulated output signals with the N bi-directional vector modulators; receiving and combining the N inbound modulated output signals with the splitter/combiner; and producing a single combined inbound RF signal with the splitter/combiner. 
   Numerous benefits may be afforded by the disclosed embodiments and include bi-directional vector modulation to allow the device to be utilized in both transmit and receive modes without introducing a differential phase shift between the incoming and outgoing signals. In addition, the vector modulator has the ability to achieve continuous phase shifts from 0 to 360 degrees (rather than discrete phase shifts), and has the ability to achieve phase shifts by adjusting attenuation. These features make the device less expensive to manufacture and ideal for phased array antenna systems when amplitude tapering is desired to optimize antenna pattern. 
   Another embodiment discloses a further benefit of the bi-directional vector modulator that has the ability to adjust attenuation to achieve phase shift. Yet, another embodiment of the present invention demonstrates benefit over prior art disclosing a phased array antenna system utilizing a bi-directional vector modulator to act in both signal transmission and reception and produce amplitude tapering of the antenna. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     In the drawings, 
       FIG. 1  is schematic drawing showing a bi-directional vector modulator. 
       FIG. 2  a vector diagram for the bi-directional vector modulator showing the magnitudes and phases of the first stage of the device illustrated in FIG.  1 . 
       FIG. 3  a vector diagram for the bi-directional vector modulator showing the magnitudes and phases of the second stage of the device illustrated in FIG.  1 . 
       FIG. 4  is a table showing the bi-directional vector modulator controls that adjust the magnitude and phase for the four quadrants of phase. 
       FIG. 5  is schematic drawing showing an additional embodiment of a bi-directional vector modulator. 
       FIG. 6  is schematic drawing showing an embodiment of a bi-directional vector modulator incorporated into a phased array antenna application. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
   While this invention is susceptible to embodiment in many different forms, there is shown in the drawings and will be described herein in detail specific embodiments thereof with the understanding that the present disclosure is to be considered as an exemplification of the principles of the invention and is not to be limited to the specific embodiments described. 
     FIG. 1  is schematic drawing showing a bi-directional vector modulator. This type of apparatus can be used in phase shifters for phased array antennas, which use the same radiating/receiving elements for both transmitting and receiving. As shown in  FIG. 1 , the signal may be traveling either from point X  102  to point Y  140  (transmission), or from point Y  140  to point X  102  (reception). 
   As can be seen in  FIG. 1 , a signal S X    104  traveling from point X  102  to point Y  140  enters a first splitter/combiner (power divider) PD 1   106 . This first splitter/combiner  106  is capable of splitting (transmission) or combining (reception) signals in phase and maintains the relationship of the output signals (i.e., when splitting, the phase shifts of the signals at the output are the same). This first splitter/combiner  106  splits signal S X    104  into signals S X1    108  and S X2    110 . Each of the split signals S X1 ,  108  and S X2    110  is input into variable attenuators VA 1   112  and VA 2   114 . Variable attenuators VA 1   112  and VA 2   114  produce attenuated output S A    116  and S B    118  respectively. In an additional embodiment of the invention, the variable attenuators  112 ,  114 ,  128  and  130  may be step attenuators with an appropriately small step size. The attenuated output S A    116  and S B    118  are input into a quadrature hybrid coupler  120  also known as a branch line coupler (BLC). This quadrature hybrid coupler  120  produces output signals S C    122  and S D′   124 . 
   The quadrature hybrid coupler  120  sums the input signals S A    116  and S B    118  to produce output signals S C    122  and S D    124  that are in quadrature, i.e., phased shifted by 90 degrees. Signal S C    122  is therefore, the superposition of signal S A    116  phase shifted by 90 degrees (lagging) and S B    118  phase shifted by 180 degrees (leading). Similarly, signal S D′   124  is the superposition of signal S A    116  phase shifted by 180 degrees (lagging) and S B    118  phase shifted by 90 degrees (leading). Thus, the amplitude of signal of the transmission output signals can be represented by:
 
Amp  S   C =1/√2[( S   A *exp( j 90°))+( S   B *exp( j 180°))]
 
Amp  S   D′ =1/√2[( S   A *exp( j 180°))+( S   B *exp( j 90°))]
 
   This quadrature hybrid coupler  120  acts identically in the reverse direction maintaining the bi-directional ability of the device. This can be demonstrated using the above argument and substituting the output signals for input signals. Therefore, the amplitude of signal of the reception output signals can be represented by:
 
Amp  S   A =1/√2[( S   C *exp( j 90°))+( S   D′ *exp( j 180°))]
 
Amp  S   B =1/√2[( S   C *exp( j 180°))+( S   D′ *exp( j* 90°))]
 
   Output signal S D′   124  is further applied to a 180 degree phase shifter  142 , which is typically a transmission line that produces a phase shifted output S D    126 . In an additional embodiment of the invention, this phase shifter  142  could also be realized by a transformer or LC ladder network. Signals S C    122  and S D    126  are transmitted through variable attenuators VA 3   128  and VA 4   130  to produce attenuated output S Y1    132  and S Y2    134  respectively. The attenuated output S Y1    132  and S Y2    134  are applied to a second splitter/combiner PD 2   136 . As with PD 1   106 , PD 2   136  is capable of combining (transmission) or splitting (reception) signals in phase and maintains the relationship of the output signals (i.e., when splitting, the phase shifts of the signals at the output are the same). 
   This second splitter/combiner  136  combines signals S Y1 ,  132  and S Y2    134  in phase to output final signal S Y    138  to point Y  140 . Phase shift and gain are adjusted by varying attenuators VA 1   112 , VA 2   114 , VA 3   128 , and VA 4   130 . 
   As mentioned above, the device of  FIG. 1  is bi-directional and can be described as a receiver with a signal traveling from point Y  140  to point X  102 . Input signal S y    138  enters a second splitter/combiner PD 2   136 . This second splitter/combiner splits signal S y    138  into signals S y1    132  and S y2    134 . Each of the split signals S y1    132  and S y2    134  is input into variable attenuators VA 3   128  and VA 4   130 . Variable attenuators VA 3   128  and VA 4   130  produce attenuated output S C    122  and S D    126  respectively. Output signal S D    126  is further applied to a 180 degree phase shifter  142 , which is typically a transmission line that produces a phase shifted output S D′   124 . 
   The attenuated output S C    122  and S D′   124  are input into a quadrature hybrid coupler  120  also known as a branch line coupler (BLC). This quadrature hybrid coupler  120  produces output signals S A    116  and S B    118 . 
   The quadrature hybrid coupler  120  sums the input signals S C    122  and S D′   124  to produce output signals S A    116  and S B    118  that are in quadrature, i.e., phased shifted by 90 degrees. Signal S A    116  is therefore, the superposition of signal S C    122  phase shifted by 90 degrees (lagging) and S D    124  phase shifted by 180 degrees (leading). Similarly, signal S B    118  is the superposition of signal S C    122  phase shifted by 180 degrees (lagging) and S D′ ,  124  phase shifted by 90 degrees (leading). 
   Signals S A    116  and S B    118  are transmitted through variable attenuators VA 1   112  and VA 2   114  to produce attenuated output S X1    108  and S X2    110  respectively. The attenuated output S X1    108  and S X2    110  are applied to a first splitter/combiner PD 1   106 . This first splitter/combiner  106  combines signals S X1    108  and S X2    110  in phase to output final signal S X    104  to point X  102 . Phase shift and gain are similarly adjusted by varying attenuators VA 1   112 , VA 2   114 , VA 3   128 , and VA 4   130 . 
   Analysis of the bi-directional vector modulator  100  disclosed in  FIG. 1  can be performed most easily by use of the principle of superposition. For example, when attenuator VA 2   114  is set to nearly infinite attenuation and attenuator VA 1   112  is set to its least attenuation (maximum signal throughput), the signal S C    122  reaching attenuator VA 3   128  is shifted 90 degrees from signal S X1    108  at the input of attenuator VA 1   112 . Since the signal at S D′   124  is 180 degrees phase shifted from signal S X1    108  at the input at attenuator VA 1   112 , signal S D    126  is 360 degrees phase shifted, i.e. it is in phase with the signal S X1    108  at the input of attenuator VA 1   112 . 
   When attenuator VA 1   112  is set to maximum attenuation (minimum signal transmission) and attenuator VA 2   114  is set to minimum attenuation, the signal at S D′   124  is 90 degrees delayed from the signal S X2    110  at the input of attenuator VA 2   114 . Therefore, the signal S D    126  is 270 degrees delayed from S X2    110  at the input of attenuator VA 2   114 . Signal S C    122  at the input of attenuator VA 3   128  is 180 degrees delayed from signal S X2    110  at the input of attenuator VA 2 . Output signals at S Y1    132  at the output of attenuator VA 3   128  and S Y2    134  at the output of attenuator VA 4  are added together as vectors by the second splitter/combiner (power divider)  136 . The first splitter/combiner  106  splits signal S X  evenly, and presents signals S X1    108  and S X2    110  with the same phase to the input ports of the quadrature coupler  120 . Therefore, the above superposition argument is validated in that the argument assumes equal phase signals. 
   In one embodiment of the invention, two of the attenuators adjust phase and one adjusts magnitude, while the fourth is set to zero transmission (i.e. infinite attenuation). In that case, the two attenuations that control phase are constrained to have a net magnitude of 1. For phase shifts of 0 to 90 degrees, for example:
 
( VA   4 ) 2 +( VA   3 ) 2 =1 
 
   In this example, the magnitude of the vector modulator is controlled by attenuator VA 1  and
 
VA 3 =sin(θ) 
 
VA 4 =cos(θ) 
 
   Similar results are obtained for the other four phase quadrants. 
   One possible embodiment of the present invention uses Wilkerson dividers for splitters PD 1   106  and PD 2   136 , but other possibilities also exist, including transformers. Similarly, the quadrature hybrid could be realized in microstrip and stripline circuits with a branch line coupler, but it could also be realized with a Lange coupler, transformer, or other reactive power divider. Attenuators VA 1   112 , VA 2   114 , VA 3   128 , and VA 4   130  could be made using any number of techniques that are insensitive to signal direction, including PIN diodes, digitally controlled attenuators, FET attenuators, etc. 
     FIG. 2  is a drawing showing a vector diagram for the bi-directional vector modulator described in FIG.  1 .  FIG. 2  illustrates the magnitudes and phases of the signals of the paths corresponding to the first stage of the device of FIG.  1 . Demonstrated in this diagram are signals S C    222  and S D′   224 , the outputs of the quadrature hybrid coupler  120 , and signal S D    226  which is a result of an additional 180 degree phase shift of S D′   224  by a phase shifter  142 . 
     FIG. 3  is a drawing showing a vector diagram for the bi-directional vector modulator in FIG.  1 .  FIG. 3  illustrates the magnitudes and phases of the signals of the paths corresponding to the second stage and the resultant output S Y    138  of the device of FIG.  1 . This stage of the device is input at the variable attenuators VA 3   128  and VA 4   130  by signals S C    122  and S D    126 . The output signals of the second stage attenuators S Y1    332  and S Y2    334  are added together as vectors by the second splitter/combiner  136 . Thus, a phase and amplitude controlled signal output S Y1    338  is obtained as a vector sum of the vectors shown in FIG.  3 . 
     FIG. 4  is a table showing the bi-directional vector modulator controls to adjust the magnitude and phase for the four quadrants of phase. By adjusting the signal amplitude of the bi-directional vector modulator of  FIG. 1 , using attenuators VA 1   112 , VA 2   114 , VA 3   128  and VA 4   130 , an arbitrary phase shift and somewhat arbitrary loss can be achieved. Besides undesired parasitic losses in the system, it can be shown that some loss is associated with any phase shift. 
   It can therefore be mathematically described in the following manner: the voltage attenuation of each attenuator VA 1   11   2 , VA 2   114 , VA 3   128  and VA 4   130  is represented by A, B, C and D, respectively. Attenuations are defined as the voltage at the output divided by the voltage at the input, and are therefore continuously valued between 0 and 1. Therefore, a signal traveling from X  102  to Y  140  as in FIG.  1 : 
       Y   =       X   2     ⁡     [     AD   -   BC   +     j   ⁢           ⁢     (       A   ⁢           ⁢   C     -   BD     )         ]           
 
   Where Y is the output voltage when the input voltage is X and where j represents the imaginary number. A, B, C and D are voltage gains through the attenuator. Because VA 1   112 , VA 2   114 , VA 3   128  and VA 4   130  are attenuators, their associated value is restricted to the continuous range from 0 to 1. Because we are dealing with a passive reciprocal network, the same equation will apply for a signal traveling from Y to X. 
   The above equation shows that both positive and negative values can be achieved for both the real and imaginary parts, by adjusting the positive real valued attenuations (A, B, C and D). The table shown in  FIG. 4  outlines the preferred approach to adjusting the magnitude and phase for the four quadrants of phase. 
   As described in  FIG. 1 , two of the attenuators adjust phase and one adjusts magnitude, while the fourth is set to zero. In that case, it is most convenient to constrain the two attenuations that control phase to have a net magnitude of 1. So, for phase shifts of 0 to 90 degrees, for example:
 
 D   2   +C   2 =1 
 
   In which case, the magnitude of the vector modulator is controlled by attenuator A and
 
 C=sin(θ)  
 
 D=cos(θ)  
 
   Similar results are obtained for the other four phase quadrants. 
     FIG. 5  is schematic drawing showing an additional embodiment of a bi-directional vector modulator. As shown in  FIG. 5 , the signal may be traveling either from point X  502  to point Y  540 , or from point Y  540  to point X  502 . As can be seen in  FIG. 5 , a signal S X    504  traveling from point X  502  to point Y  540  enters a 180-degree rat race coupler RC 1   506 . This rat race coupler RC 1   506  splits signal S X    504  into signals S X1    508  and S X2    510  and maintains the phase relationship of the output signals. Each of the split signals S X1    508  and S X2    510  is applied to variable attenuators VA 1   512  and VA 2   514 . Variable attenuators VA 1   512  and VA 2   514  produce attenuated output S A    516  and S B    518  respectively. In an additional embodiment of the invention, the variable attenuators  512 ,  514 ,  528  and  530  may be step attenuators with an appropriately small step size. The attenuated outputs S A    516  and S B    518  are transmitted through a quadrature hybrid coupler  520  that produces output signals S C    522  and S D′   524 . Signals S C    522  and S D    526  are applied to variable attenuators VA 3   528  and VA 4   530  to produce attenuated output S Y1    532  and S Y2    534  respectively. The attenuated output S Y1    532  and S Y2    534  are transmitted through a second 180-degree rat race coupler RC 2   536  to output final signal S Y    538  to point Y  540 . Phase shift and gain are adjusted by varying attenuators VA 1   512 , VA 2   514 , VA 3   528 , and VA 4   530 . 
   A similar analysis of the bi-directional vector modulator  500  can be performed in a similar manner by use of superposition as was described for the bi-directional vector modulator  100  disclosed in FIG.  1 . 
   The device of  FIG. 5  is bi-directional and can be described as a receiver with a signal traveling from point Y  540  to point X  502 . Input signal S Y    538  enters a second 180-degree rat race coupler RC 2   536 . This second coupler  536  splits signal S Y    538  into signals S Y1    532  and S Y2    534 . Each of the split signals S Y1    532  and S Y2    534  is input into variable attenuators VA 3   528  and VA 4   530 . Variable attenuators VA 3   528  and VA 4   530  produce attenuated output S C    522  and S D    526  respectively. The attenuated output S C    522  and S D    526  are input into a quadrature hybrid coupler  520  also known as a branch line coupler (BLC). This quadrature hybrid coupler  520  produces output signals S A    516  and S B    518 . 
   The quadrature hybrid coupler  520  sums the input signals S C    522  and S D    526  to produce output signals S A    516  and S B    518  that are in quadrature, i.e., phased shifted by 90 degrees. Signal S A    516  is therefore, the superposition of signal S C    522  phase shifted by 90 degrees (lagging) and S D    526  phase shifted by 180 degrees (leading). Similarly, signal S B    518  is the superposition of signal S C    522  phase shifted by 180 degrees (lagging) and S D    526  phase shifted by 90 degrees (leading). 
   Signals S A    516  and S B    518  are transmitted through variable attenuators VA 1   512  and VA 2   514  to produce attenuated output S X1    508  and S X2    510  respectively. The attenuated output S X1    508  and S X2    510  are applied to a first 180-degree rat race coupler RC 1   506 . This first coupler  506  combines signals S X1    508  and S X2    510  in phase to output final signal S X    504  to point X  502 . Phase shift and gain are similarly adjusted by varying attenuators VA 1   512 , VA 2   514 , VA 3   528 , and VA 4   530 . 
     FIG. 6  is schematic drawing showing an embodiment of a bi-directional vector modulator incorporated into a phased array antenna application  600 . This diagram represents a complete phased array antenna system with N elements and does not address the physical relationship between the N elements, which is one of the things that determines antenna performance. The antenna array can be used to broadcast an RF wave in a transmit mode and then receive reflected portions of the broadcast wave in a receive mode as in a radar applications, or an independent transmission can be received in the receive mode as in other communication applications. Those skilled in the art may design a phased array antenna system finding various ways that the elements could be arranged within an antenna system while keeping within the spirit and intent of the present invention. These might include a circular array, a rectangular array, a linear array, a planar array, or the like or any combination thereof. 
   To transmit a radio frequency wave, the system is set to transmit mode where a signal originates from a radio transceiver  602  and is split through the N:1 splitter/combiner  604 , which equally divides power and applies it to all N outputs  610 ,  620  and  630 . This N is typically, but not limited to, some power of two, i.e.,  16 ,  32 ,  64 ,  128 ,  256 , etc. The split output signals are transmitted to a corresponding number bi-directional vector modulators  612 ,  622  and  632 , as detailed in  FIG. 1  or FIG.  5 . Because the N:1 splitter/combiner  604  and the bi-directional vector modulators  612 ,  622  and  632  introduces loss, a bi-directional amplifier  614 ,  624  and  634  can be placed on the output of each bi-directional vector modulators  612 ,  622  and  632  to compensate for this loss. This amplified signal is then transmitted to the radiating/receiving element  616 ,  626  and  636  that produces an outbound radio frequency wave. Again, the radiating/receiving element  616 ,  626  and  636  is an additional item that those skilled in the art could realize with numerous device types, while keeping within the spirit and intent of the present invention. These radiating/receiving elements  616 ,  626  and  636  could be for example, a slot wave guide, a radiating patch, microstrip patch, a monopole element, a dipole element or the like. 
   To receive a radio frequency wave, the system is set to receive mode where a signal is received by a radiating/receiving (receiving) element  616 ,  626  and  636 . The signal is amplified to compensate for internal signal loss and propagated to the bi-directional vector modulators  612 ,  622  and  632  and then to the N:1 splitter/combiner (combiner)  604  and back to the radio transceiver  602 . 
   One of the basic principles of antenna design is called the theorem of reciprocity. It states that antennas transmitting and antennas receiving perform the same. When you measure an antenna transmitting, you will get an identical result as an antenna receiving. This means that if the antenna were to receive (the bi-directional amp and system switch set to receive mode), then the antenna pattern would be exactly the same when receiving as transmitting. This is beneficial because it is easier to measure an antenna when it is transmitting then when it is receiving, because the signal to noise ratio is much higher in the transmit mode. 
   For instance, if a plain wave were incident upon the elements (in whatever pattern they may be aligned), the bi-directional vector modulator can be adjusted such that only the radiation from a particular direction constructively interferes with each other in all N elements, and radiation from other directions destructively interferes. Therefore, the signal can be greatly enhanced by carving constructive interference. 
   The controller  606  is connected to each of the bi-directional vector modulators  612 ,  622  and  632  to adjust the four attenuator values (A, B, C and D detailed in  FIG. 4 ) for all N elements. Each of the N elements has four attenuator values, and the controller  606  is adjusting those four attenuator values in order to point in a particular direction. In the circumstance where the antenna needs to point in a different direction when receiving versus transmitting, the attenuator values will change. In the situation where the controller  606  does not need to change control parameters whether transmitting or receiving, the attenuator values do not change. This demonstrates one benefit of the current invention. Once the antenna has been aligned for transmission, the antenna holds that alignment during reception (the converse is also true). If the antenna needs to communicate with a number of varied locations, the controller can change the antenna direction at will with a single circuit for transmitting and receiving. 
   The foregoing description of the invention has been presented for purposes of illustration and description. It is not intended to be exhaustive or to limit the invention to the precise form disclosed, and other modifications and variations may be possible in light of the above teachings. The embodiment was chosen and described in order to best explain the principles of the invention and its practical application to thereby enable others skilled in the art to best utilize the invention in various embodiments and various modifications as are suited to the particular use contemplated. It is intended that the appended claims be construed to include other alternative embodiments of the invention except insofar as limited by the prior art.