Abstract:
A demodulator for use with a digital wireless communication system is disclosed, that comprises a DC offset controller for removing a DC offset of a modulated signal that is input to the demodulator, a complex multiplying unit for complex-multiplying an output signal of the DC offset controller, a phase detector for detecting an amplitude error signal and a phase error signal from an output signal of the complex multiplying unit, an LPF (low pass filter) for outputting a low band component of the phase error signal, and an NCO (numerical controlled oscillator) for converting an output signal of the LPF into a sin component and a cos component that have orthogonal relation, wherein the sin component and the cos component are input to the complex multiplying unit, and wherein the amplitude error signal, the sin component, and the cos component are input to the DC offset controller.

Description:
BACKGROUND OF THE INVENTION 
     1. Field of the Invention 
     The present invention relates to a demodulator for used with a wireless communication receiver, in particular, to a demodulator having a DC offset controller that controls a frequency offset with an angle signal and an error signal. 
     2. Description of the Related Art 
     In recent years, digital wireless communication systems using microwaves and ultra-microwaves have been widely used along with analog wireless communication systems. In particular, mobile wireless communication systems such as portable telephone system and PHS (Persona Handyphone System) have been actively developed and invested as an infrastructure of digital communication lines. 
     In a demodulator of a digital wireless communication system, a frequency offset has been controlled so as to suppress data errors and accurately demodulate data. FIG. 11 is a block diagram showing the structure of a demodulator for use with a conventional digital wireless communication system. In FIG. 11, the demodulator comprises multiplying units  1  and  2 , an oscillator  5 , A/D converters  3  and  4 , a complex multiplying unit  6 , a phase detector  7 , an LPF  8 , an NCO (Numerical Controlled Oscillator)  9 , a DC offset controller  11 , and a Π/2 shifter  12 . 
     In a receiver of a digital wireless communication system, a received radio signal is down-converted into a desired IF signal. The resultant IF signal is supplied to the demodulator as shown in FIG.  11 . The detecting method of the demodulator is for example quasi-synchronous detecting method. The input modulated signal is an orthogonally modulated signal corresponding to QPSK method or QAM method. Orthogonal components (channels) of the modulated signal are denoted by Ich (In-Phase Channel) and Qch (Quadri-Phase Channel). 
     The oscillator  5  is a local oscillator with the same frequency as that of the input IF signal. An output signal of the oscillator  5  and a signal of which the phase of the output signal of the oscillator  5  is shifted for Π/2 by the Π/2 shifter  12  are multiplies by the IF signal of the IF-IN. Thus, signal components I and Q that are base band signals of the signals Ich and Qch are obtained. The signal components I and Q are supplied to the A/D converters  3  and  4 , respectively. The A/D converters  3  and  4  convert the signal components I and Q into digital signals I 1  and Q 1 , respectively. Since the demodulator performs the quasi-synchronous detection, the digital signals I 1  and Q 1  are not perfect base band signals. Instead, the digital signals I 1  and Q 1  contain carrier frequencies. 
     The DC offset controller  11  inputs the digital signals I 1  and Q 1 , removes DC offset components from the digital signals I 1  and Q 1 , and outputs signals I 2  and Q 2 . 
     The complex multiplying unit  6  inputs the signals I 2  and Q 2 , removes carrier frequencies from the signals I 2  and Q 2  using rotation angle information sin and cos that are input from the NCO  9 , and outputs resultant signals Ich 4  and Qch 4 . 
     The phase detector  7  inputs the signals Ich 4  and Qch 4  from the complex multiplying unit  6  and outputs a phase error signal Pd 1 . The phase error signal Pd 1  is supplied to the LPF  8 . The LPF  8  smooths the phase error signal Pd 1  and outputs the resultant phase error signal Pd 2  to the NCO  9 . 
     The NCO  9  converts the phase error signal Pd 2  that is input from the LPF  8  into rotation angle signals sin and cos. 
     The output signals Ich 4  and Qch 4  of the complex multiplying unit  6  are converted into a serial signal. Thus, original digital data can be obtained. 
     Next, another structure of the DC offset controller  11  will be described. In the structure, the DC offset controller  11  controls a DC offset with information other than error information that is output from the complex multiplying unit  6 . 
     FIG. 7 shows an influence of a DC offset upstream of the complex multiplying unit that compensates a frequency/phase offset. In FIG. 7, a polarity determining unit  121  determines the polarities of digital signals Ich 1  and Qch 1 . An adding unit  123  adds the output signals of the polarity determining unit  121  with an output signal of the adding unit  123 . A flip-flop F/F  122  temporarily stores an output signal of the adding unit  123 . Corresponding to the stored polarity, the polarity determining unit  121  determines whether or not a DC offset is present. 
     FIG. 12 shows an influence of a DC offset upstream of the complex multiplying unit  6  that compensates a frequency/phase offset in a modulating method of which signal points are present on a concentric circle. Dots on the circle represent signals in the case that DC offset components are present on both Ich and Qch. Since the frequence/phase offset has not been compensated, the signals are offset to the origin of the coordinates. When the DC offset is removed from the signals, the resultant signals are represented as a solid circle whose center is at the origin of the coordinates. 
     FIG. 13 shows a signal on only Qch (or Ich) in the case that the signal has a DC offset. The signal reciprocally moves on a straight line. The signal deviates to the axis I. In FIG. 13, the probability of which the value on Q′ch is positive is higher than the probability of which the value on Q′ch is negative. When the DC offset is removed from the signal, the resultant signal distributes symmetrically to the axis I as denoted by a solid line Qch. 
     FIG. 14 is a block diagram showing an example of the structure of the conventional DC offset controller  11 . In FIG. 14, the DC offset controller  11  comprises adding units  111  and  112  and LPFs  113  and  114 . In the DC offset controllers  11 , the adding units  111  and  112  remove low band components including DC components that are output components of the LPFs  113  and  114  and output signals Ich 2  and Qch 2  of which DC offset components are removed. 
     In the conventional structure, although error information is not obtained, when DC offset components are present, since the probability of which each of signals Ich and Qch is positive is different from the probability of which each of signals Ich and Qch is negative, the DC offset components can be controlled corresponding to polarity signals on Ich and Qch. Since the polarity signals are used as control signals, LPFs shown in FIG. 7 or  9  are used. When an input signal is positive, a subtracting operation or a count-down operation is performed. When an input signal is negative, an adding operation or a count-up operation is performed. Thus, the probability of which each signal is positive becomes the same as the probability of which each signal is negative. Thus, the DC offset components are removed. 
     However, as described above, in the conventional structure, although DC offset components can be removed, since only polarities as control information are used, the DC offset control becomes coarse. 
     As denoted by I-Q coordinates shown in FIG. 15, when signals that are output from the complex multiplying unit  6  contain DC offset components, the signals are denoted as circles with radiuses of DC offset components. FIG. 15 shows signals in the QPSK modulating method. Likewise, in other modulating methods, signals are denoted as circles with centers of correct signal points. Thus, when the distance between signal points is short, the signals are easily affected by noise and thereby the error rate characteristics deteriorate. 
     In particular, when a signal is transmitted with many value data for a large capacity of a communication line, due to a DC offset, the distance between signal points further shortens. Thus, the error rate characteristics remarkably deteriorate. 
     SUMMARY OF THE INVENTION 
     An object of the present invention is to provide a demodulator for accurately removing a DC offset so as to more suppress deterioration of error rate characteristics than conventional demodulators in the case that a modulating method with many-value data and short signal distance is used. 
     A first aspect of the present invention is a demodulator for use with a digital wireless communication system, comprising a DC offset controller for removing a DC offset of a modulated signal that is input to the demodulator, a complex multiplying unit for complex-multiplying an output signal of the DC offset controller, a phase detector for detecting an amplitude error signal and a phase error signal from an output signal of the complex multiplying unit, an LPF (low pass filter) for outputting a low band component of the phase error signal, and an NCO (numerical controlled oscillator) for converting an output signal of the LPF into a sin component and a cos component that have orthogonal relation, wherein the sin component and the cos component are input to the complex multiplying unit, and wherein the amplitude error signal, the sin component, and the cos component are input to the DC offset controller. 
     The phase detector multiplies and adds an output signal of the complex multiplying unit and the amplitude error signal that is output from the error detector and obtains the phase error signal. 
     The DC offset controller multiplies the amplitude error signal by the sin component, multiplies the amplitude error signal by the cos component, adds one of the multiplied results and the other of the multiplied results, subtracts one of the multiplied results from the other of the multiplied results, extracts low band components of the added result and the subtracted result through the LPF, adds or subtracts the low band components to/from the orthogonal modulated signal component, and outputs the result as an orthogonal modulated signal component. 
     A second aspect of the present invention is a digital wireless communication receiver having a demodulator for demodulating transmitted digital data, wherein the demodulator is the demodulator of the first aspect. 
     These and other objects, features and advantages of the present invention will become more apparent in light of the following detailed description of a best mode embodiment thereof, as illustrated in the accompanying drawings. 
    
    
     BRIEF DESCRIPTION OF DRAWINGS 
     FIG. 1 is a block diagram showing the structure of a demodulator according to a first embodiment of the present invention; 
     FIG. 2 is a block diagram showing a phase detector according to the first embodiment of the present invention; 
     FIG. 3 is a block diagram showing the structure of an LPF according to the first embodiment of the present invention; 
     FIG. 4 is a block diagram showing the structure of an NCO according to the first embodiment of the present invention; 
     FIG. 5 is a block diagram showing the structure of a complex multiplying unit according to the first embodiment of the present invention; 
     FIG. 6 is a block diagram showing the structure of a DC offset controller according to the first embodiment of the present invention; 
     FIG. 7 is a block diagram showing the structure of an LPF according to the first embodiment of the present invention; 
     FIG. 8 is a graph showing I-Q characteristics of an error signal corresponding to an error signal according to the first embodiment of the present invention; 
     FIG. 9 is a block diagram showing the structure of an LPF according to a second embodiment of the present invention; 
     FIG. 10 is a block diagram showing the structure of a DC offset controller according to a third embodiment of the present invention; 
     FIG. 11 is a block diagram showing the structure of a conventional demodulator; 
     FIG. 12 is a graph showing I-Q characteristics of a conventional DC offset control; 
     FIG. 13 is a graph showing I-Q characteristics of a conventional DC offset control; 
     FIG. 14 is a block diagram showing the structure of a conventional DC offset controller; and 
     FIG. 15 is a graph showing I-Q characteristics of a conventional DC offset control. 
    
    
     DESCRIPTION OF PREFERRED EMBODIMENTS 
     Next, with reference to the accompanying drawings, embodiments of the present invention will be described. 
     [First Embodiment] 
     Next, with reference to FIG. 1, a first embodiment of the present invention will be described. FIG. 1 shows the structure of a demodulator according to the first embodiment of the present invention. The demodulator comprises multiplying units  1  and  2 , an oscillator  5 , A/D converters  3  and  4 , a complex multiplying unit  6 , a phase detector  7 , an LPF  8 , an NCO (Numerical Controlled Oscillator)  9 , a DC offset controller  10 , and a Π/2 shifter  12 . 
     In the following description, the detecting method of the demodulator is for example quasi-synchronous detecting method. The input modulated signal is an orthogonally modulated signal corresponding to QPSK method or QAM method. Orthogonal components (channels) of the modulated signal are denoted by Ich and Qch. 
     An input signal IF-IN of the demodulator shown in FIG. 1 is obtained from an IF amplifying circuit (that amplifies an intermediate frequency signal) through an antenna (that receives a radio signal), a radio frequency amplifying circuit (that amplifies the received signal), a mixing circuit (that converts the amplified signal into an intermediate frequency signal), and a local oscillation circuit (that inputs a local oscillation frequency signal to the mixing circuit). Next, each circuit block shown in FIG. 1 will be described in detail. 
     The oscillator  5  is a local oscillator that generates a local oscillation signal at a frequency that is almost the same as that of the input IF signal. An output signal of the oscillator  5  and a signal of which the phase of the output signal of the oscillator  5  is shifted for Π/2 by the Π/2 shifter  12  are multiplied by the IF-IN. Thus, signal components Ich and Qch are obtained. 
     The A/D converters  3  and  4  convert the signal components on Ich and Qch into digital signals Ich 1  and Qch 1 . Since the demodulator performs the quasi-synchronous detection, the digital signals Ich 1  and Qch 1  are not perfect base band signals. Instead, the digital signals Ich 1  and Qch 1  contain carrier frequencies. 
     The DC offset controller  10  inputs the digital signals Ich 1  and Qch 1 , removes DC offset components from the digital signals Ich 1  and Qch 1  using amplitude error signals Ei and Eq that are input from the phase detector  7  and rotation angle signals sin and cos that are input from the NCO  9 , and outputs signals Ich 2  and Qch 2 . 
     The complex multiplying unit  6  inputs the signals Ich 2  and Qch 2 , removes carrier frequencies from the signals Ich 1  and Qch 1  using rotation angle information sin and cos that are input from the NCO  9 , and outputs resultant signals Ich 3  and Qch 3 . 
     Next, the phase detector  7  outputs a phase error signal Pd and amplitude error signals Ei and Eq using the signals Ich 3  and Qch 3  that are input from the complex multiplying unit  6 . Thus, carrier frequencies have been removed from the signals Ich 3  and Qch 3 . Next, the phase error signal Pd 1  is supplied to the LPF  8 . The LPF  8  smooths the phase error signal Pd 1  and outputs the resultant signal as a phase error signal Pd 2  to the NCO  9 . 
     The NCO  9  converts the phase error signal Pd 2  that is input form the LPF  8  into rotation angle signals sin and cos. 
     The signals Ich 3  and Qch 3  are converted into serial signals I and Q. The signals I and Q are output to a data signal processing circuit. 
     Next, each circuit block will be described in detail. 
     FIG. 2 is a block diagram showing an example of the structure of the phase detector  7 . In FIG. 2, the phase detector  7  comprises error detectors  71  and  72 , multiplying units  73  and  74 , and an adding unit  75 . 
     The error detectors  71  and  72  detect errors of the input signals Ich 3  and Qch 3  against their normal signal point positions and output the detected errors as amplitude error signals Ei and Eq. In this example, when the input signals Ich 3  and Qch 3  deviate in the positive direction against the normal signal point positions, the error signals Ei and Eq become negative. In contrast, when the input signals Ich 3  and Qch 3  deviate in the positive direction against the normal signal point positions, the error signals Ei and Eq become positive. With the error signals Ei and Eq and polarity signals Di and Dq of the signals Ich 3  and Qch 3  (the polarity signals Di and Dq are MSBs of the signals Ich 3  and Qch 3 ), a phase error signal Pd 1  is obtained. 
     FIG. 3 is a block diagram showing an example of the structure of the LPF  8 . In FIG. 3, the LPF  8  comprises a flip-flop F/F  81 , multiplying units  82  and  83 , and adding units  84  and  85 . The multiplying units  82  and  83  multiply the input phase error signal Pd 1  by multiplying coefficients α and β, respectively. The multiplying coefficients α and β are parameters that define characteristics of the LPF  8 . The multiplying coefficients α and β are optimized so as to satisfy the follow-up characteristics against a required input signal and stability of an output signal. 
     The LPF  8  is a conventional quadratic lag lead filter. Normally, in a carrier reproduction loop composed of the complex multiplying unit  6 , the phase detector  7 , the LPF  8 , and the NCO  9 , the LPF  8  should be structured at least quadratically. To do that, with two or three stages of the LPFs  8 , a high order low pass filter is structured. Thus, with a high order carrier reproduction loop, a frequency offset can be canceled. 
     FIG. 4 is a block diagram showing an example of the structure of the NCO  9 . In FIG. 4, the NCO  9  comprises a integrating unit  93 , a cos ( )  91 , and a sin ( )  92 . The integrating unit  93  integrates the phase error signal Pd 2  smoothed by the LPF  8  and outputs a frequency error signal θ. The cos ( )  91  and the sin ( )  92  are calculating units that input the frequency error signal θ and output cos (θ) and sin (θ) as angle signals sin and cos. 
     FIG. 5 is a block diagram showing an example of the structure of the complex multiplying unit  6 . In FIG. 5, the complex multiplying unit  6  comprises multiplying units  61  to  64  and adding units  65  and  66 . 
     The multiplying units  61  and  64  multiply output signals Ich 2  and Qch 2  of the DC offset controller  10  by the output signal cos (θ) of the NCO  9 . In addition, the multiplying units  62  and  63  multiply the signals Ich 2  and Qch 2  by the output signal sin (θ) of the NCO  9 . The adding unit  65  subtracts an output signal of the multiplying unit  63  from an output signal of the multiplying unit  61  and outputs a signal Ich 3 . The adding unit  64  adds an output signal of the multiplying unit  62  and an output signal of the multiplying unit  64   a  and outputs a signal Qch 3 . 
     The complex multiplying unit  6  shown in FIG. 5 removes carrier frequencies and phase offset components from the signals Ich 2  and Qch 2 . 
     FIG. 6 is a block diagram showing the structure of the DC offset controller  10 . In FIG. 6, the DC offset controller  10  comprises adding units  101  and  102 , multiplying units  107  to  110 , adding units  105  and  106 , and LPFs  103  and  104 . 
     In FIG. 6, the multiplying units  109  and  110  multiply the output error signals Ei and Eq of the phase detector  7  by the output signal cos (θ) of the NCO  9 , respectively. The multiplying units  107  and  108  multiply the error signals Ei and Eq by the output signal sin (θ) of the NCO  9 , respectively. The adding unit  105  adds an output signal of the multiplying unit  109  and an output signal of the multiplying unit  108 . The adding unit  106  subtracts an output signal of the multiplying unit  107  from an output signal of the multiplying unit  110 . Output signals of the adding units  105  and  106  are input to the LPF  103  and the LPF  104 . The LPF  103  and the LPF  104  pass low band components of the output signals of the adding units  105  and  106 . The output signals Ich 1  and Qch 1  of the A/D converters  3  and  4  are added and output as signals Ich 2  and Qch 2 , respectively. 
     The calculating circuit composed of the adding units  105  and  106  and the multiplying units  107  to  110  in FIG. 6 are almost the same as the complex multiplying unit  6  shown in FIG.  5 . The rotating direction of the calculating circuit shown in FIG. 6 is different from the rotating direction of the complex multiplying unit  6  shown in FIG.  6 . The error signals Ei and Eq that are input from the phase detector  7  are inversely rotated with the angle signals sin and cos. The complex multiplying unit  6  restores the original signal. The inversely rotated signals are smoothed by the LPFs  103  and  104 . The resultant signals are added to the input signals Ich 1  and Qch 1 . Thus, the DC offset components are removed. 
     FIG. 7 is a block diagram showing an example of the structure of each of the LPFs  103  and  104  of the DC offset controller  10 . In FIG. 7, each of the LPFs  103  and  104  comprises a polarity determining unit  121 , a flip-flop F/F  122 , and an adding unit  123 . Normally, a DC offset is a regular offset due to imperfectness of hardware. Thus, the LPFs  103  and  104  can be accomplished by simple filters that integrate a polarity signal (namely, the LSB of a signal). 
     [Operation of First Embodiment] 
     Next, the operation of the DC offset controller  1  that is a feature of the present invention will be described as the operation of the first embodiment. 
     In FIG. 1, the DC offset controller  10  is operated with the amplitude error signals Ei and Eq before the complex multiplying unit  6  rotates the phases of the signals. However, the amplitude error signals Ei and Eq are obtained from signals whose phases have been rotated. Thus, error information of which the phases of signals have not been rotated should be estimated with the obtained error information. 
     Thus, the amplitude error signals Ei and Eq and the phase error signals Pd 1  are obtained by the phase detector  7  shown in FIG.  2 . The amplitude error signals Ei and Eq can be obtained by detecting errors of the input signals Ich 3  and Qch 3  against the original signal point positions. With the amplitude error signals Ei and Eq and the polarity signals Di and Dq (MSBs) of the output signals Ich 3  and Qch 3  of the complex multiplying unit  6 , the phase error signal Pd 1  is expressed by the following formula. 
     
       
           Pd   1 = Ei·Dq−Eq·Di   
       
     
     A phase error is converted into a frequency error by the NCO  9  shown in FIG.  4 . Since a frequency is obtained by integrating a phase, after the phase error signal Pd 1  is smoothed by the LPF  8  and then integrated, a frequency error signal θ is obtained. The frequency error signal θ is converted into rotation angle signals sin (θ) and cos (θ). The rotation angle signals sin (θ) and cos (θ) are output from the NCO  9 . FIG. 5 is a block diagram showing an example of the structure of the complex multiplying unit  6 . The block diagram shown in FIG. 5 represents calculations expressed by the following formulas. 
     
       
           Ich   3 = Ich   2  cos(θ)− Qch   2  sin(θ) 
       
     
     
       
           Qch   3 = Ich   2  sin(θ)+ Qch   2  cos(θ) 
       
     
     In other words, calculations for rotating the signals Ich 2  and Qch 2  by the angle signal θ obtained from the error signal are repeated in the loop composed of the phase detector  7 , the LPF  8 , and the NCO  9 . Thus, the carrier frequencies and phase offset components contained in the signals Ich 1  and Qch 1  are controlled. 
     On the other hand, calculations performed by the multiplying units and the adding units of the DC offset controller  10  shown in FIG. 6 are expressed by the following formulas. 
     
       
           Ei′=Ei  cos(θ)− Eq  sin(θ) 
       
     
     
       
           Eq′=Ei  sin(θ)+ Eq  cos(θ) 
       
     
     Thus, the inverse rotation of the complex multiplying unit  6  is performed for the signals Ei and Eq. 
     FIG. 8 shows the relation between the error signals E and E′ on I-Q phase plane. A vector that orients from the correct signal point position to the output signal of the complex multiplying unit  6  is an error signal obtained by the phase detector  7 . When the phase of the error signal is rotated for θ by the complex multiplying unit  6 , the input signal (whose phase has not been rotated) of the complex multiplying unit  6  is obtained by rotating the output signal of the complex multiplying unit  6  for −θ. Thus, the error signal E′ whose phase has not been rotated is obtained by rotating the error signal E for −θ. As shown in FIG. 6, when the error signal E′ is smoothed by the LPFs  103  and  104  and input to the input signals Ich 1  and Qch 1 , since the error signal converges to zero, the DC offset components can be removed. 
     Thus, the DC offset components can be removed by the DC offset controller  10  using the error signals Ei and Eq and the phase error signals sin and cos detected by the phase detector  7 . The frequency offset can be removed from the AC error signal by the carrier reproduction loop composed of the complex multiplying unit  6  and the NCO  9 . 
     In other words, in the demodulator, the DC offset controller  10  complex-multiplies the error signals Ei and Eq that are output from the phase detector  7  by sin and cos that are output from the NCO  9  so as to estimate error information whose phases have not been rotated using the error signals Ei and Eq whose phases have been rotated by the complex multiplying unit  6 . The DC offset controller controls (removes) the DC offset components with the estimated error information. 
     Since the amplitude error signals Ei and Eq contains accurate DC offset information, the demodulator according to the present invention can more accurately control (remove) the DC offset components than the conventional demodulator that controls them with information that has not been complex-multiplied. 
     In the first embodiment, the detecting method of the demodulator is the quasi-synchronous detecting method and the input modulation signal is a signal that is orthogonally modulated corresponding to the QPSK or QAM method. However, as long as error information whose frequencies/phase offset components have been compensated and rotation angle information whose frequencies/phase offset components are compensated are obtained, the detecting method of the demodulator may not be the quasi-synchronous detecting method. In other words, another modulating method such as PSK or APSK may be used. 
     [Second Embodiment] 
     Next, with reference to FIG. 9, a second embodiment of the present invention will be described. FIG. 9 is a block diagram showing a second example of the structure of each of the LPFs  103  and  104  of the DC offset controller  10 . In FIG. 9, each of the LPFs  103  and  104  comprises a polarity determining unit  124  and an up-down counter U/D  125 . 
     The polarity determining unit  124  represents only two states of positive and negative unlike with the polarity determining unit  121  shown in FIG.  7 . Thus, the polarity determining unit  124  outputs only one bit. The counter U/D  125  performs a count-down operation when the input signal of the polarity determining unit  124  is positive. The counter U/D  125  performs a count-up operation when the input signal of the polarity determining unit  124  is negative. The structure shown in FIG. 9 is simpler than the structure shown in FIG.  7 . Thus, in the structure shown in FIG. 9, the circuit scale can be reduced. 
     In addition, each of the LPFs  103  and  104  may be a quadratic LPF. In this case, although the circuit scale slightly becomes large, the follow-up characteristics and initial response characteristics due to a fluctuation of the DC offset components can be improved. 
     [Third Embodiment] 
     Next, with reference to FIG. 10, a third embodiment of the present invention will be described. FIG. 3 is a block diagram showing the structure of a DC offset controller  10  according to a third embodiment of the present invention. In the DC offset controller  10  shown in FIG. 10, the complex multiplying unit  6  shown in FIG. 1 is used. In addition, a polarity inverter  1012  is used. The polarity inverter  1012  inverts the polarity of the signal sin and thereby inverts the rotation direction. When a desired signal is represented with a 2&#39;s complement, after all bits are inverted, “1” should be added. However, when an error of one LSB is permitted, all bits can be inverted. Thus, when a selector is connected to input/output of the complex multiplying unit  6  and the complex multiplying unit is shared on time division basis, the circuit scale can be reduced. 
     Thus, according to the present invention, with an error signal against a normal signal position, DC offset control can be performed so that the error signal becomes zero. Consequently, DC offset components can be more accurately removed. Thus, in the case that a modulating method with many value data and short signal distance is used, the error rate characteristics are further improved. 
     Although the present invention has been shown and described with respect to a best mode embodiment thereof, it should be understood by those skilled in the art that the foregoing and various other changes, omissions, and additions in the form and detail thereof may be made therein without departing from the spirit and scope of the present invention.