Abstract:
A system and method is disclosed that provides a digital self-adjusting power supply for semiconductor digital circuits. The power supply provides a substantially constant minimum supply voltage with regard to process corner, junction temperature, external voltage source, load variation, and operating frequency. The system comprises a slack time detector, a voltage adjuster, and a digital pulse width modulation (PWM) modulator. The system supplies a minimum required voltage without the used of a band gap or reference voltage. A finite state machine is also used to minimize oscillations introduced by start-up, load transients, frequency changes, and the like, thereby eliminating the need for a proportional integrator differentiator (PID) circuit.

Description:
TECHNICAL FIELD OF THE INVENTION 
   The present invention is generally directed to manufacturing technology for semiconductor devices and, in particular, to digital self-adjusting power supply systems that provide a substantially constant minimum power supply voltage. 
   BACKGROUND OF THE INVENTION 
   Semiconductor process technology continues to reach lower voltages and deeper sub-micron sizes. As the number of transistors per integrated circuit chip has continued to increase, two critical circuit design issues have been presented. The first design issue is the non-uniformity of process parameters within a single die. The second design issue is the increment in power consumption per die. 
   In deep sub-micron circuit design, variations due to the non-uniformity of process parameters within a single die cause differences in transistor and interconnect characteristics across the single die. These differences in turn negatively impact the performance of the circuit because they generate deviations in MOSFET (Metal Oxide Semiconductor Field Effect Transistor) drive current. This results in propagation delay distributions of the critical path across a chip. 
   Furthermore, the distribution of process parameters expands from die to die within a single wafer as well as within a lot. After fabrication, operating variations such as power supply voltage, chip temperature, and across-chip temperature also affect the magnitude of the propagation delay. By combining both operational and process induced variations, the magnitude of the propagation delay fluctuates from eighteen percent (18%) to thirty two percent (32%). 
   The yield of CMOS (Complimentary Metal Oxide Semiconductor) logic circuits that satisfy a specific performance requirement is significantly influenced by the magnitude of critical path delay deviations due to both operational and intrinsic parameter fluctuations. There are two approaches to compensating for the impact of these parameter fluctuations and achieving a desired yield. The first approach is to reduce the performance by operating at a lower clock frequency. The second approach is to increase the supply voltage. 
   In many portable computing devices such as MP-3 players and digital cameras, the full processing power of a processor is not required all the time. There are certain times when an operating frequency can be reduced. A lower frequency corresponds to a longer allowable delay. This longer time margin also allows a supply voltage level to be lowered whereas the applied lower voltage increases the propagation delay. 
   Because power consumption is quadratic with the supply voltage and proportional to operating frequency, reducing both operating frequency and supply voltage allows an excellent energy efficient operation. This technique is referred to as adaptive voltage scaling (AVS). Adaptive voltage scaling decreases power consumption without sacrificing performance provided that the tasks that are performed are finished within an allowed time. From the trade-off between performance and energy consumption, supplying just enough voltage to a system at a given frequency represents its optimum power consumption. 
   Although the operating frequency limits allowable propagation delay, this delay strongly depends on intrinsic process parameters, supply voltage and junction temperature (referred to by the abbreviation “PVT”). The propagation delay in a MOSFET is proportional to the product of the active resistance of the MOSFET (designated R ON ) and the load capacitance (designated C L ). The resistance R ON  is given by: 
   
     
       
         
           
             
               
                 
                   R 
                   ON 
                 
                 = 
                 
                   
                     V 
                     DD 
                   
                   
                     
                       β 
                       ⁡ 
                       
                         ( 
                         
                           
                             V 
                             DD 
                           
                           - 
                           
                             V 
                             T 
                           
                         
                         ) 
                       
                     
                     α 
                   
                 
               
             
             
               
                 ( 
                 1 
                 ) 
               
             
           
         
       
     
   
   where α is the velocity saturation term, β is the process transconductance parameter, V DD  is the supply voltage, and V T  is the threshold voltage. The load capacitance C L  is given by:
 
 C   L   =C   D   +C   G   +C   W   (2)
 
   where C D  is the drain capacitance, C G  is the gate capacitance, and C W  is the interconnect capacitance. 
   Process parameters and operating junction temperature are not controllable but the supply voltage is controllable. Therefore, if the supply voltage can be adjusted to guarantee the same propagation delay regardless of the other operating conditions, then various simulations are not needed to assure proper functionality. Instead, only one simulation (i.e., the worst case simulation with a small margin) is needed to guarantee proper operation after fabricating the design. 
   If a design is fabricated as the best process corner, and is operating at low temperature, it needs to be less than three fourths of the minimum supply voltage required to operate at the worst case simulation. This results in power savings by reducing the power supply voltage with regard to process and temperature. 
   On the other hand, by dynamically adjusting the supply voltage, an individual die is adjusted to a desirable performance. Typically the supply voltage should be raised for dies that are operating slowly and lowered for dies that are operating quickly. In this manner the yield is dramatically improved by the adaptive voltage scaling (AVS) system. 
   Typical prior art digital control loop circuits employ direct current (DC) to direct current (DC) switching converter circuits. A DC-to-DC switching converter circuit is a device that converts a first DC voltage to a second DC voltage with very little power loss. A boost converter is one type of DC-to-DC switching converter that receives an input voltage of one polarity and outputs a voltage of the same polarity that is larger than the input voltage. A buck converter is another type of DC-to-DC switching converter that receives an input voltage of one polarity, and outputs a voltage of the same polarity that is smaller than the input voltage. 
     FIG. 1A  shows a block diagram of an embodiment of an exemplary prior art buck converter  100 . As shown in  FIG. 1A , converter  100  has a power supply circuit  102  that receives a pulse width modulated signal PWS and generates a supply voltage VDD in response to the pulse width modulated signal PWS. 
   Power supply circuit  102  comprises a DC voltage source  110  and an n-channel MOS transistor  112  that has a source, a drain connected to voltage source  110 , and a gate connected to receive the pulse width modulated signal PWS. Power supply circuit  102  also comprises a diode D that has an input connected to ground, and an output connected to the source of transistor  112 . 
   Power supply circuit  102  also comprises an inductor L and a capacitor C that form an LC network. Inductor L has a first end connected to the source of transistor  112 , and a second end connected to a supply node NS. Capacitor C has a first end connected to the supply node NS and a second end connected to ground. In addition, a load  114  is connected between the supply node NS and ground. 
   In operation, the pulse width modulated signal PWS turns transistor  112  on and off, which outputs a pulsed current to the LC network. The LC network averages the pulsed current to generate the supply voltage VDD on the supply node NS. The supply voltage VDD has a value that is approximately equal to the duty cycle of the pulse width modulated signal PWS multiplied times the voltage of voltage source  110 . For example, when a fifty percent (50%) duty cycle signal is used with a two volt (2 V) voltage source, a supply voltage VDD of approximately one volt (1 V) results. 
   Diode D is used to provide a continuous conductive path for inductor L. When transistor  112  turns on, current is driven into inductor L, which stores the energy. When transistor  112  turns off, the stored energy is transferred to capacitor C as the voltage on the input of inductor L drops below ground. When the voltage on the input of inductor L drops below ground, diode D turns on, thereby providing a continuous conductive path. 
   Referring again to  FIG. 1A , converter  100  also comprises a compensation circuit  116  that adjusts the duty cycle of the pulse width modulated signal to maintain a roughly constant supply voltage VDD in response to changes in process, voltage, and temperature (PVT). 
   As shown in  FIG. 1A , compensation circuit  116  comprises an adjuster block  120  that divides down (or gains up) the supply voltage VDD on the supply node NS to output an adjusted voltage VA. In addition, compensation circuit  116  comprises an error block  122  that compares the adjusted voltage VA to a band gap reference voltage VBG to generate an error voltage VER. 
   Compensation circuit  116  further comprises a proportional integrator differentiator (PID)  124  that responds to changes in the error voltage VER, and outputs a control voltage VC in response thereto, and a pulse width modulator  126  that outputs the pulse width modulated signal PWS. 
   The LC network is a two pole resonant system that stores energy. As a result, any transient introduced to the system, such as a start up transient or a load transient, causes a disturbance. The disturbance causes a response through the LC network that produces an oscillation or a ringing effect. 
   PID  124  eliminates the ringing effect by providing a zero that cancels one of the poles in the LC network. Canceling one of the poles, in turn, results in a first order system that is inherently stable. In this manner PID  214  converts the second order response of the LC network to a first order response. 
   Pulse width modulator  126  varies the positive widths (duty cycles) of the pulses in the pulse width modulated signal PWS in response to the control voltage VC output by PID  214 . For example, a centered control voltage produces a pulse width modulated signal PWS with a fifty percent (50%) duty cycle. In addition, a driver block  128  drives the pulse width modulated signal PWS onto the gate of transistor  112 . 
   In operation, the band gap reference voltage VBG is generated by a band gap circuit, and is roughly constant over changes in process, voltage, and temperature (PVT). Under normal operating conditions, the adjusted voltage VA and the band gap reference voltage VBG are equal and the error voltage VER is zero. A zero error voltage produces a control voltage VC that, in turn, produces a pulse width modulated signal PWS. 
   When the adjusted voltage VA varies due to changes in PVT, error block  122  outputs the error voltage VER with a non-zero value. PID  124  filters the error voltage VER to output a control voltage VC. Pulse width modulator  126  responds to the control voltage VC by varying the duty cycle of the pulse width modulated signal PWS which, in turn, changes the supply voltage VDD. This process continues until the adjusted voltage VA and the band gap voltage VBG are again equal. 
   Thus, compensation circuit  116  adjusts the duty cycle of the pulse width modulated signal PWS to maintain a roughly constant supply voltage VDD in response to changes in process, voltage, and temperature (PVT). 
   One drawback of switching converter  100  is that diode D introduces an undesirable resistance. One approach to reducing the resistance introduced by diode D is to use a synchronous rectifier.  FIG. 1B  shows a block diagram that illustrates a typical prior art synchronous rectifier  150 . 
   As shown in  FIG. 1B , synchronous rectifier  150  comprises a PMOS driver transistor  152  that has a source connected to a supply voltage VDD, a drain connected to inductor L through an inductor node NL, and a gate. Synchronous rectifier  150  also comprises a NMOS driver transistor  154  that has a source connected to ground, a drain connected to inductor L through the inductor node NL, and a gate. 
   As also shown in  FIG. 1B , synchronous rectifier  150  also comprises a gate signal generator  156  that receives a pulse width modulated signal PWS, and outputs non-overlapping gate signals G 1  and G 2  to PMOS transistor  152  and NMOS transistor  154 , respectively. 
   In operation, when the pulse width modulated signal PWS transitions low, generator  156  turns off NMOS transistor  154  via gate signal G 2 , and then turns on PMOS transistor  152  via gate signal G 1 . When PMOS transistor  152  turns on, transistor  152  sources current into inductor node NL. 
   When the pulse width modulated signal PWS transitions high, generator  156  turns off PMOS transistor  152  via gate signal G 1 , and then turns on NMOS transistor  154  via gate signal G 2 . When NMOS transistor  154  turns on, transistor  154  provides a path to ground to provide a continuous conductive path at the inductor node NL for an inductor. 
   Another drawback of switching converter  100  is that the converter requires a band gap reference voltage source to respond to changes in PVT. A band gap reference voltage source, however, is a complex circuit that provides only a roughly constant voltage over changes in PVT. 
   A further drawback of switching converter  100  is that switching converter  100  requires a proportional integrator differentiator (PID), which is also a complex circuit, in order to compensate for the second order effects of the LC network. The implementation of a PID circuit adds a heavy hardware burden because a PID control law is typically expressed as a current and prior values of duty ratio, and error, and includes coefficients that require evaluation using multiplication. 
   In view of the disadvantages inherent in the prior art switching converter circuitry, it would be advantageous to have an improved system and method for compensating for the impact of operational and intrinsic parameter fluctuations on circuit performance by adaptively adjusting the supply voltage to guarantee proper operation of digital processors. It would be advantageous to have an improved system and method for providing a DC-to-DC switching converter that provides a substantially constant supply voltage over changes in PVT without employing a band gap reference voltage source and without employing a PID circuit. 
   It would also be advantageous to have an improved system and method for digitally self-adjusting a minimum power supply system to regulate a power supply voltage to a minimum value required to operate at a given PVT and a given frequency. 
   Before undertaking the Detailed Description of the Invention below, it may be advantageous to set forth definitions of certain words and phrases used throughout this patent document: the terms “include” and “comprise,” as well as derivatives thereof, mean inclusion without limitation; the term “or,” is inclusive, meaning and/or; the phrases “associated with” and “associated therewith,” as well as derivatives thereof, may mean to include, be included within, interconnect with, contain, be contained within, connect to or with, couple to or with, be communicable with, cooperate with, interleave, juxtapose, be proximate to, be bound to or with, have, have a property of, or the like; and the term “controller” means any device, system or part thereof that controls at least one operation, such a device may be implemented in hardware, firmware or software, or some combination of at least two of the same. It should be noted that the functionality associated with any particular controller may be centralized or distributed, whether locally or remotely. Definitions for certain words and phrases are provided throughout this patent document, those of ordinary skill in the art should understand that in many, if not most instances, such definitions apply to prior uses, as well as future uses, of such defined words and phrases. 

   
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention and its advantages, reference is now made to the following description taken in conjunction with the accompanying drawings, in which like reference numerals represent like parts: 
       FIG. 1A  is a block diagram illustrating a prior art buck converter  100 ; 
       FIG. 1B  is a block diagram illustrating a prior art synchronous rectifier  150 ; 
       FIG. 2  is a block diagram illustrating a digital self adjusting minimum power supply system in accordance with the principles of the present invention; 
       FIG. 3  is a circuit diagram illustrating a slack time detector of the present invention; 
       FIG. 4  is a circuit diagram of an exemplary delay cell of the slack time detector shown in  FIG. 3 ; 
       FIG. 5  is a timing diagram of delayline outputs and clock signals of the present invention; 
       FIG. 6  is a block diagram illustrating a voltage adjuster of the present invention; 
       FIG. 7  is a block diagram illustrating a one zero edge detector and an illustrative example of an encoding procedure of the present invention; 
       FIG. 8  is a block diagram illustrating a frequency compensator of the present invention; 
       FIG. 9  is a block diagram illustrating a PVT compensator of the present invention; 
       FIG. 10  is a diagram illustrating supply voltages corresponding to equivalent propagation delay for a best case condition, a typical case condition, and a worst case condition; 
       FIG. 11  is a circuit diagram illustrating a clock generator of the present invention; 
       FIG. 12  is a timing diagram of clock signals of the clock generator of the present invention; 
       FIG. 13  is a state diagram illustrating the states of the finite state machine of the present invention; 
       FIG. 14  is a diagram illustrating lock acquisitions of the supply voltage in accordance with the principles of the present invention; 
       FIG. 15  is a block diagram illustrating a pulse width modulation (PWM) modulator of the present invention; 
       FIG. 16  is a diagram illustrating measured output voltage as a function of tap value in accordance with the principles of the present invention; 
       FIG. 17  is a diagram illustrating how a tap selector of the invention chooses four tap signals from sixteen tap signals of a slack time detector. 
   

   DETAILED DESCRIPTION OF THE INVENTION 
     FIGS. 2 through 17 , discussed below, and the various embodiments used to describe the principles of the present invention in this patent document are by way of illustration only and should not be construed in any way to limit the scope of the invention. Those skilled in the art will understand that the principles of the present invention may be implemented in any type of suitably arranged electronic device. 
     FIG. 2  is a block diagram illustrating a digital self adjusting minimum power supply system  200  in accordance with the principles of the present invention. The system  200  comprises a closed loop controller  210  that receives a fixed frequency clock signal CLK and frequency information FI(5:0), a DC-DC buck converter  220 , and a processor  230  as a load current source. The closed loop controller  210  comprises a slack time detector  240 , a voltage adjuster  250 , and a pulse width modulation (PWM) modulator  260 . 
   The closed loop controller  210  performs the following discrete-time compensation:
 
 D ( n+ 1)= F ( n )+α E ( n )+ A ( n )  (3)
 
   The expression D(n+1) represents the next value of the duty ratio. The expression F(n) represents the frequency compensation factor. The expression E(n) represents the detected error. The expression A(n) represents the accumulated compensation. The expression alpha (α) represents an error scaling factor. 
   In indicated the expression in Equation (3), the control approach does not request any previous values, although an accumulated compensation factor A(n) is needed. The variations of PVT and load are dynamically updated in the compensation factor. During updating, the duty ratio is controlled by a finite state machine (FSM) that maintains a substantially constant supply voltage. The error scaling factor alpha (α) is used to increase the error resolution and it is implemented by a right shift function instead of multiplication. 
   The ability to determine the minimum voltage required for a given operating frequency needs to continuously monitor a critical path delay through the digital circuitry with respect to PVT, load, and frequency. This is referred to as a slack time detection. The slack time detector  240  of the present invention comprises a chain of delay cells (DL 1  through DL 27 ), and a tap register (TS 12  through TS 27 ) as shown in  FIG. 3 . A circuit diagram of an exemplary delay cell of the slack time detector  240  is shown in  FIG. 4 . 
   The principle of a delayline as an analog to digital converter (ADC) is based on the relation between a supply voltage and the propagation delay. The slack time detector  240  determines the voltage level of the delayline&#39;s supply according to the propagation delay through the delayline. In other words, the delay along with the supply voltage is converted to a digital value by sampling the delayline. 
   There are three considerations for designing a delayline to monitor the critical path timing over PVT variation. The first requirement is to avoid the worst case cross-over effect on non-linear characteristics between delay and voltage. The cross-over is worse when the delay becomes longer or the voltage becomes lower. Therefore, the margin for proper circuit operation should be applied for high cross-over ratio. 
   The second requirement is to determine the resolution of the delayline. A fine step size can result in very slow settling. On the other hand, a coarse step size can cause hysteretic oscillation. The third requirement is to minimize the hardware burden for the delayline. As semiconductor fabrication technology improves, the circuit delay is shorter, and in turn the number of needed cells to implement a critical path delay is larger. 
   From the third delayline design requirement, the cells of the delayline should have low performance. A NOR structure slowly transits at a high-to-low transition compared to a NAND structure. However, low-to-high transition time of a NOR gate is the same as that of a NOT gate. Therefore, a pair of NOR and NOT gates is selected as the unit delay cell. 
   The number of delay cells between taps is determined so that the accumulated steps voltage by increasing a step voltage makes one more tap active than the prior accumulated steps voltage at the worst case. The number of delay cells between taps varies because the propagation delay is not a linear function of the supply voltage. 
   The inverted input of the delay cell (NOR-INV) receives the input clock signal or the output of the delay cell, and the non-inverted input is connected to a delayline enable signal. The sampling clock signal is lagged from the input clock signal by one quarter (¼) of a period. The tap register samples the values of the delayline at one quarter (¼) period intervals after the input clock pulse has begun propagating through the delayline.  FIG. 5  shows the timing of the inputs DX 12  through DX 27  at each tap, the input clock signal ICLK, the sample clock signal SCLK, and the delayline enable signal RSTN. 
   The magnitude of the supply voltage is inferred by determining how far along the delayline the input clock pulse propagates in one fourth (¼) of a period. Therefore, a delayline in the negative feedback path of a closed loop reflects variations in circuit performance in response to variations of PVT, load and frequency and adaptively scales the regulated voltage of a buck converter via a loop controller. 
   The delayline is characterized at the worst case with regard to fixed frequency input sources. Delay of the delayline implies its process corner, junction temperature, and supply voltage at a given workload. From this measurement, a desirable constant supply voltage in response to variations of PVT and load can be determined to guarantee the propagation delay just less than the critical path delay limitation that assures proper operation. 
   As shown in  FIG. 6 , the voltage adjuster  250  comprises a datapath block  610  and a control block  615 . Datapath block  610  comprises an error compensator  620 , a frequency compensator  625 , and a process, voltage, and temperature (PVT) compensator  630 . The major role of the voltage adjuster  250  is to compensate a supply voltage error at a given frequency from the measurement of the slack time detector  240  and to provide a desirable constant voltage level against variations of frequency as well as PVT. In addition, for high-speed and low-overshoot/undershoot start-up, it controls soft-start operation. 
   Error Compensator. The role of the error compensator  620  is to detect the voltage error E(n) (in Equation (3)), and to generate a proportionally compensated value. It receives the propagation delay word TX(27:12) from the slack time detector  240  and detects the position of one and zero pair of taps as shown in  FIG. 7 . The error compensator  620  converts the propagation delay position to an error voltage E(n) by comparing it to the reference delay position along with supply voltage at the worst case, and in turn generates a proportionally compensated delay word ECW(5:0) that represents a reference value at a default frequency plus a compensated error value. 
   Frequency Compensator. The frequency compensator  625  adjusts the duty cycle of the pulse width modulated (PWM) pulse according to the desirable supply voltage in response to a given frequency. A more detailed view of the frequency compensator  625  is shown in  FIG. 8 . The first subtractor SUBL 1  in  FIG. 8  generates a difference between the frequency information FI(5:0) and the internal reference voltage level RFI(5:0). The difference implies the desirable voltage variation in response to a given frequency. 
   The up/down counter CNT receives the difference, and counts up or down at the load signal LOAD until the output of the counter equals the difference. This prevents the desirable supply level along with the frequency variation from abruptly changing to reduce the ringing effect. The second subtractor SUB 2  receives the proportionally compensated propagation delay word ECW(5:0) from the error compensator  620  and the shift-left counter number for subtrahend (from the left shift circuit LSH), and generates a frequency compensated propagation delay word FCW 1 (5:0). 
   The compensated error-step from the delayline is the same as the resolution of the delayline which is six (6) bits. However, the resolution of the digital-to-analog converter (DAC) should be higher than that of the analog-to-digital converter (ADC), and the error step should have higher resolution. To increase the control resolution, the proportionally compensated error-value should be scaled close to the resolution of the DAC. 
   The third subtractor SUB 3  provides a reference supply voltage word FCW 2 (5:0) at a given frequency FI(5:0). The one bit higher resolution compensated word FCW(7:1) is generated by adding the frequency compensated word to the reference supply voltage word. However, the seven (7) bit digital-to-analog converter (DAC) for the six (6) bit analog-to-digital converter (ADC) is not enough to avoid limit-cycling. 
   Therefore, dither logic generates a least significant bit (LSB) of the frequency compensated word FCW( 0 ). The LSB is toggled (average value) unless the desirable voltage is not achieved. When the supply voltage reaches a target voltage, it is set to zero. The frequency compensated word represents the reference value plus half of the compensated error value, and an additional least significant bit (LSB). The frequency compensated word FCW( 0 ) corresponds to the frequency compensated factor F(n) in Equation (3). 
   Process, Voltage, and Temperature (PVT) Compensator. The process, voltage, and temperature (PVT) compensator  630  comprises an internal dynamic voltage reference source MX 1 , a pulse width generator ADD 1 , and a ringing stopper MX 2  as shown in  FIG. 9 . The internal dynamic voltage reference source MX 1  adds or subtracts one, two, or three steps according to the one (1) step, two (2) steps, or three (3) steps increment or decrement indicators (U 1 , U 2 , U 3 , D 1 , D 2 , D 3 ) and generates an internal dynamic voltage reference IREF(7:0). 
   The reference value compensates the fluctuations due to process and temperature variations as well as the quantization error of external supply voltage.  FIG. 10  illustrates the equivalent supply voltages that ensure the same propagation delay at different operational and intrinsic parameters.  FIG. 10A  illustrates a supply voltage corresponding to a best case.  FIG. 10B  illustrates a supply voltage corresponding to a typical case.  FIG. 10C  illustrates a supply voltage corresponding to a worst case. 
   The pulse width generator ADD 2  receives the frequency compensated word FCM(7:0) and the accumulated compensation IREF(7:0). The accumulated compensation IREF(7:0) corresponds to the accumulated compensation factor A(n) in Equation (3). The pulse width generator ADD 2  generates a normal PWM pulse width NPW(7:0). 
   The ringing stopper MX 2  receives three inputs. They are the shift-left PWM pulse width (from the left shift circuit LSH), the normal PWM pulse width, and the shift-righted PWM pulse width (from the right shift circuit RSH). The ringing stopper MX 2  outputs a pulse width word PW(7:0) in response to selection signals UP, NR, and DOWN. Because the high-valued derivative direction of supply voltage during frequency-changing or starting-up is not changed by only step-size compensated value NPW(7:0), the emphasized activation (double or half size of the PWM pulse) is needed. 
   Control Block of Voltage Adjuster. As shown in  FIG. 6 , the control block  615  comprises a clock generator  635 , a tap selector  640 , a finite state machine (FSM)  645 , a control signal generator  650 , and a false low level detector  655 . 
     FIG. 11  is a circuit diagram illustrating an advantageous embodiment of the clock generator  635 .  FIG. 12  is a timing diagram of the clock signals of the clock generator  635 . The clock generator  635  outputs a one quarter (¼) frequency input clock signal ICLK and a one quarter (¼) period lagged sample clock signal SCLK from an external clock input CLK. The clock generator  635  also outputs a one thirty second ( 1/32) frequency PWM loading signal LOAD and a delayline reset signal RSTN from the sample clock signal SCLK. The clock generator  635  also outputs a frequency doubled clock DCLK from the external input clock CLK. 
   As shown in  FIG. 17 , in response to external frequency information FI(5:0) the tap selector  640  chooses four (4) taps (such as prior-tap INCSET, center-tap CNTTAP, next-tap DECSET, and next next-tap DECSET 2 ) from the sixteen (16) taps (TX 12 -TX 27 ) of the slack time detector  240 . The control signal generator  650  outputs increment or decrement indicators UP, UP 3 , DOWN, and DOWN 3  in response to the status of the four (4) taps of the tap detector, the state CS 2 -CS 0  from the Finite State Machine (FSM)  645 , and external frequency information. 
   The Finite State Machine (FSM)  645  is shown in more detail in  FIG. 13 . The Finite State Machine (FSM) receives monitor signals from the other control modules and the datapath block  610 , and outputs control signals to the datapath block  610 . The monitor signals comprise the first tap TX 12 , the last tap TX 27 , the four (4) taps INCSET, CNTTAP, DECSET, and DECSET 2 , two frequency change indicators (INC and DEC), and the external frequency information FI(5:0). The control signals comprise a current state CS(2:0), a low voltage state signal LOW, a high voltage state signal HIGH, and a clear signal CLR of a PWM pulse width counter. 
   The states from 000 to 011 control a soft-start routine to avoid large overshoot/undershoot with high-speed saturation. In the states from 100 to 111, the FSM  645  controls a duty cycle of the digital pulse width modulation (DPWM) modulator  260  according to the current voltage status detected by the value of the four (4) taps that varies over PVT conditions. The outputs of the FSM  645  are used to change the counter number of the PVT compensator at weighted steps and to double or half of the counter number. This accelerates the supply voltage to be stable fast over various value of PVT in start-up as well as in normal operation. 
   The DAC&#39;s higher resolution than that of the ADC gives multiple DAC values for a given ADC output. During a high to low voltage transition along with frequency change, the highest voltage among the DAC level at the same ADC level may be a settling point. The false low-level detector  655  steps down the voltage level until a true minimum DAC level according to a given frequency is detected. 
   As shown in  FIG. 14 , the supply voltage settles down at the first high DAC level in an ADC bin by the operation of the FSM  645  (Coarse Lock Acquisition), and in turn converges to the bottom DAC level by the operation of the false low level detector  655  (Fine Lock Acquisition). 
   DPWM Pulse Modulator.  FIG. 15  illustrates a more detailed view of the DPWM Pulse Modulator  260 . The DPWM Pulse Modulator  260  comprises a loadable down counter  1510  and a pulse generator  1520 . The down counter  1510  loads the PWM pulse width PW(7:0) from the voltage adjuster  250  by the load signal LOAD, changing a binary output of down counter  1510  as a count in response to the frequency doubled clock signal DCLK. DPWM Pulse Modulator  260  outputs a pulse modulated signal PWM defined by the binary input value PW(7:0) of down counter  1510 . 
   The single loop design of the present invention eliminates most of the analog circuit including references, ramp generators, and ADC comparators that are usually associated with analog synchronous buck converters. A driver chip that is less than one millimeter (1 mm) on a side can easily be manufactured and placed in a low cost SOT23-5 or a μSMD bump chip package. 
   A one half micron (0.5 μm) power CMOS process serves as an excellent choice for the driver for power supplies less than five and one half volts (5.5 V) which is the case in virtually all present day cell phones. Additional features such as over and under voltage protection, thermal shutdown, and dead time (non overlapping phase) can also be included. The driver can be made with an arbitrarily large current (e.g., one ampere) without significantly impacting stability and efficiency. The driver chip can also be directly driven from a low voltage source because level shifters are included in the input. In addition, the need for trims and voltage corrections that are normally associated with analog switcher circuits are eliminated. 
   The digital self-adjusting minimum power supply system of the present invention comprises a driver  270  having an input connected to the DPWM Modulator  260 . As shown in  FIG. 2 , driver  270  drives an LC filter. The LC filter generates a DC supply voltage on adaptive voltage scaling (AVS) Power Supply Node  280  in response to a pulse modulated signal PMS. Because the LC filter is an intermittently unstable system, it is very important to choose proper values for the inductance L and the capacitance C. 
   From the maximum ripple requirement, the LC product can be chosen from Equation (4): 
   
     
       
         
           
             
               
                 
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                     ⁢ 
                     LC 
                   
                 
               
             
             
               
                 ( 
                 4 
                 ) 
               
             
           
         
       
     
   
   The expression ΔV represents the ripple voltage. The expression Vin represents the input voltage. The letter D represents the duty ratio. The expression f SW  represents the switching frequency. The expression LC represents the product of the inductance L and the capacitance C. 
   Consider the following numerical example. If the output voltage Vout is 1.2 volts and the input voltage Vin is 3.6 volts, and if the ripple voltage ΔV is 5.0 millivolts, and the switching frequency f is six hundred twenty five kiloHertz (625 kHz), then the required value of the LC product is fifty one and two tenths microseconds squared (51.2 μsec 2 ). 
   The minimum oscillation (damped natural) frequency is expressed in Equation (5) as follows: 
   
     
       
         
           
             
               
                 
                   ω 
                   d 
                 
                 = 
                 
                   
                     
                       1 
                       LC 
                     
                     - 
                     
                       
                         R 
                         2 
                       
                       
                         4 
                         ⁢ 
                         
                           L 
                           2 
                         
                       
                     
                   
                 
               
             
             
               
                 ( 
                 5 
                 ) 
               
             
           
         
       
     
   
   The expression ωd represents the damped natural frequency. The inductance is L and the capacitance is C. The resistance R is the sum of the resistances of switch TR and ESR. From Equation (5) the minimum ωd condition is expressed in Equation (6) as follows: 
   
     
       
         
           
             
               
                 
                   
                     2 
                     R 
                   
                   ⁢ 
                   
                     
                       L 
                       C 
                     
                   
                 
                 = 
                 1 
               
             
             
               
                 ( 
                 6 
                 ) 
               
             
           
         
       
     
   
   The widths of the Field Effect Transistor (FET) drivers are sized so that the resistance R is one ohm (1Ω) in Equation (1). Therefore, the numerical value of capacitance C is approximately four (4) times the numerical value of inductance L. Using the value LC=51.2 μsec 2  one obtains an approximate value of inductance L=3.58 microHenries (μH) and an approximate value of capacitance C=14.31 microFarads (μF). 
   The digital self-adjusting minimum power supply system of the present invention provides a closed loop automatic supply adjustment mechanism to generate the optimum operating voltage for core processors in response to variations in PVT as well as those of frequency and load. 
   The switching frequency of the buffer is six hundred twenty five megahertz (625 MHz). The Digital Pulse Width Modulation (DPWM) resolution is eight (8) bits, ½ 8  and the output voltage resolution is approximately twelve and one half millivolts (12.5 mV). The frequency information from APC has a six (6) bit resolution (fifty millivolts (50 mV) at worst case) to prevent output voltage level overlapping by ringing to due ripple and noise. 
   Therefore, the output voltage of the present AVS converter can be regulated by any voltages at ADC resolution between seven tenths of a volt (0.7 V) and one and two tenths volt (1.2 V). The minimum peak ripple is five millivolts (5 mV). The power dissipation from the AVS controller is approximately one hundred microwatts (100 μW) at eight (8) bits DAC resolution. 
     FIG. 16  is a diagram illustrating measured output voltages as a function of tap value in accordance with the principles of the present invention.  FIG. 16  shows the measured output voltages at different processes corners.  FIG. 16  illustrates the equivalent tap voltages to guarantee the same propagation delay at different corners of a thirteen hundredths micron (0.13 μm) process technology. The bottom lines on the graph represent the best cases at room temperature. The middle lines on the graph represent to worst cases at room temperature. The top dashed line in the graph represents an ideal case under the worst condition. 
   The digital self-adjusting minimum power supply system of the present invention provides a substantially constant minimum supply voltage that guarantees just less propagation delay than critical path delay over changes in process, voltage and temperature (PVT), load, and frequency. The fully digital technique of the present invention provides an improved controller for AVS regulation in digital applications that present a hostile environment for noise-sensitive analog circuits. 
   The foregoing description has outlined in detail the features and technical advantages of the present invention so that persons who are skilled in the art may understand the advantages of the invention. Persons who are skilled in the art should appreciate that they may readily use the conception and the specific embodiment of the invention that is disclosed as a basis for modifying or designing other structures for carrying out the same purposes of the present invention. Persons who are skilled in the art should also realize that such equivalent constructions do not depart from the spirit and scope of the invention in its broadest form. 
   Although the present invention has been described with an exemplary embodiment, various changes and modifications may be suggested to one skilled in the art. It is intended that the present invention encompass such changes and modifications as fall within the scope of the appended claims.