Abstract:
Active load modulation antennas for contactless systems typically require the presence of a battery power source in the transponder device. The transponder typically cannot be powered by the reader device alone and also transmit an active load modulation signal. Embodiments in accordance with the invention are disclosed that allow transponder devices to transmit an active load modulation signal when powered only by the reader in the contactless system.

Description:
BACKGROUND OF THE INVENTION 
       [0001]    To guarantee interoperability between contactless card readers and transponders, international standards specify the properties of the air interface. For example, ISO/IEC 14443 is the fundamental international standard for proximity cards, ISO/IEC 10373-6 is the test standard for proximity systems, EMVCo is the industry standard for payment and ECMA 340 is the Near Field Communication (NFC) interface and protocol. Conformance of the contactless card readers and transponders to these standards is typically essential and in some instances needs to be certified by an accredited test laboratory. A number of properties are specified for the air interface of contactless products by the international standards. One property is the so-called Load Modulation Amplitude (LMA). 
         [0002]    For example, in the communication link from a device in card mode (hereinafter referred to as the transponder device) to a device in contactless reader mode (hereinafter referred to as the contactless reader), the information is communicated using load modulation. Due to the inductive proximity coupling between the loop antenna circuit of the reader and the loop antenna circuit of the transponder device, the presence of the transponder device affects the contactless reader and is typically referred to as the “card loading effect”. From the perspective of the contactless reader, a change in resonance frequency and a decrease in the Quality (Q) factor of the resonant circuit occurs. If the contactless reader/transponder device coupling system is viewed as a transformer, the transponder device represents a load to the contactless reader. Modulating the frequency and Q of the transponder loop antenna circuit produces a modulation of the load on the contactless reader. The contactless reader detects this load modulation at the reader antenna as an AC voltage. For systems compliant with ISO/IEC 14443, for example, the load modulation is applied to a sub-carrier frequency (e.g. 0.8475 MHz) of the 13.56 MHz carrier frequency specified by the standard or the 13.56 carrier frequency is directly modulated by the encoded signal for systems compliant with FeliCa, a contactless RFID smartcard system developed by Sony in Japan. 
         [0003]    Each standard typically specifies a minimum limit for the load modulation amplitude that needs to be achieved by the transponder device in card mode. 
         [0004]    Typically, restrictions such as available space or cost place strict limits on the antenna size. Furthermore, the presence of other components in close proximity to the contactless reader antenna circuit or transponder device antenna circuit effect the antenna circuit resonance properties, typically producing a shift in resonance frequency and decreasing the Q-factor. To address this issue, typically ferrite materials such as sintered or polymer ferrite foils are used for one layer of the construction of transponder and reader antennas. For example, see US Patent Publication 201100068178 A1 incorporated by reference herein. 
         [0005]    For transponder devices that are powered only by the contactless reader device, there is typically a physical limitation on the load modulation that may be achieved using conventional methods such as passive switching of a resistor or capacitor to modulate the frequency or Q-factor of the antenna resonance circuit. The physical limitation typically depends on antenna size of the transponder device, the coupling between transponder and reader, the Q-factor of the resonant circuit, the switching time and other parameters. Note, the switching time is fixed for the 847.5 kHz subcarrier frequency in context of the ISO/IEC 14443 standard. These physical limitations allow the generation of a limit curve for the minimum antenna area that can achieve compliance with the minimum load modulation specified by the standards. 
         [0006]    The minimum load modulation required can be achieved using a smaller planar loop antenna if the card mode communication is transmitted actively into the contactless reader antenna. Options exist which can induce the same voltage into the contactless reader antenna as is possible using conventional passive amplitude load modulation. For example, one option is to transmit a 13.56 MHz carrier signal that is modulated by the 847.5 kHz subcarrier frequency which is in turn modulated using the encoded data operating in card mode. 
         [0007]    However, for active load modulation to work, the active load modulation of the transponder device typically needs to be in phase with the, for example, 13.56 MHz alternating magnetic field emitted by the contactless reader. The contactless reader typically provides the time reference for communication using the contactless interface. Typical transponder devices derive the clock frequency from the exemplary 13.56 MHz carrier signal provided by the contactless reader. Therefore, the signal typically used for the communication link from the transponder device to the contactless reader is in phase with the carrier signal emitted by the contactless reader. For a transponder device actively emitting in card mode with only one antenna, however, it is typically not possible to obtain the time reference from the contactless reader carrier signal. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWING 
         [0008]      FIG. 1   a  shows active load modulation in accordance with the invention. 
           [0009]      FIG. 1   b  shows an embodiment in accordance with the invention. 
           [0010]      FIG. 1   c  shows an embodiment in accordance with the invention. 
           [0011]      FIG. 1   d  shows an embodiment in accordance with the invention. 
           [0012]      FIG. 2   a  shows the H-field for circular loop antenna. 
           [0013]      FIG. 2   b  shows induced voltage as a function of antenna overlap in accordance with the invention. 
           [0014]      FIGS. 3   a - h  shows the separate layers of an embodiment in accordance with the invention in top view. 
           [0015]      FIG. 4  shows the layers of an embodiment in accordance with the invention in side view. 
           [0016]      FIG. 5   a  shows the contours of the H-field in cross-sectional plane perpendicular to an embodiment in accordance with the invention. 
           [0017]      FIG. 5   b  shows the contours of the H-field in cross-sectional plane perpendicular to an embodiment in accordance with the invention. 
       
    
    
     DETAILED DESCRIPTION 
       [0018]    In accordance with the invention, a special antenna geometry (e.g. a planar loop, but three dimensional embodiments are also possible) together with a special receiver and driver allow a transponder device to receive the exemplary 13.56 MHz signal from the contactless reader at the same time as the transponder device is transmitting in active card mode. This allows synchronization of the active load modulation signal with the carrier signal transmitted by a contactless reader (not shown) as is shown in  FIG. 1   a  for an exemplary carrier frequency of 13.56 MHz and subcarrier frequency of 847.5 kHz. Active load modulation signal  160  uses the logical AND of synchronous carrier wave  165  with subcarrier wave  175  AND baseband signal  185  which employs Manchester coding (e.g. see  FIG. 1   a ). A carrier wave at the exemplary frequency of 13.56 MHz is actively transmitted by the contactless reader (not shown) to the transponder device (not shown). Active load modulation signal  160  is emitted from the transponder device and has the same phase relationship in every burst with synchronous carrier wave  165  provided by the contactless reader. Synchronous carrier wave  165  defines the time reference for communications between the transponder and the contactless reader. For comparison,  FIG. 1   a  also shows typical passive load modulation signal  195  at the transponder antenna. 
         [0019]      FIG. 1   b  shows an embodiment in accordance with the invention where planar loop antenna  110  comprises two individual planar coils  115  and  125 . Planar coils  115  and  125  are connected at pad  150  and shifted laterally with respect to each other so that there is nearly zero electromagnetic coupling between coils  115  and  125 . Planar coils  115  and  125  are positioned on opposite sides of substrate  120  which may be, for example, polyethylene terephthalate (PET) foil or polyvinyl chloride (PVC) foil. Planar loop antenna  110  on substrate  120  is typically placed over ferrite foil  128 . Note that ferrite foil  128  extends distance  129  beyond the last turn of coils  115  and  125 . This typically improves the performance (e.g. increased communication distance and/or allows higher bit rates) of planar loop antenna  110 . For an exemplary embodiment of planar loop antenna  110  in accordance with the invention, the dimensions of planar loop antenna  110  are about 30 mm by about 17 mm, where distance  129  is set to about 5 mm and the width of conductors  101  is about 0.4 mm (which is also the spacing between conductors  101 ). Antenna overlap  155  is the overlap between coils  115  and  125  and is about 5 mm in length for an embodiment in accordance with the invention. 
         [0020]      FIGS. 1   c  and  1   d  show two geometrical options for planar loop antenna  110  for an embodiment in accordance with the invention. Other geometrical shapes are possible as well for embodiments in accordance with the invention. Planar loop antenna  111  in  FIG. 1   c  has a circular geometry with coils  116  and  126 . Note the overlapping area between coil  116  and coil  126  and common ground  149  to which both coil  116  and coil  126  are connected. Planar loop antenna  112  has a triangular geometry with coils  117  and  127 . Note the overlapping area between coil  117  and coil  127  and common ground  148  to which both coil  117  and coil  127  are connected. 
         [0021]    The size for planar loop antenna  110  typically depends on the contactless performance that is desired. For interoperability with products that meet the ISO/IEC14443 standard, geometric size classes are defined. Typically, the largest size is the card format which is specified in ISO/IEC7810 as the ID-1 format which is about 86 mm by about 55 mm. For certain applications, the size may need to be considerably smaller, typically the smallest size would be about 5 mm by about 5 mm in accordance with the invention. 
         [0022]    Typically, the width of conductors  101  of coils  115  and  125  is in the rang of about 0.1 mm to about 3 mm for embodiments in accordance with the invention. For typical commercial processes, 0.1 mm is the lower limit on the width resolution. For etching processes, some copper thicknesses are typical. Typically 35 μm, 18 μm and 12 μm are commercially available thicknesses for conductors  110  using an etching process. Electroplating or galvanic processes allow thicknesses on the order of about 1 μm. Thickness is also dependent on the design requirements for the environment where planar loop antenna  110  will be used. 
         [0023]    The amount of current typically flowing in conductors  101  of coils  115  and  125  typically requires a certain conductor volume to avoid thermally overloading conductors  101 . Typical currents in conductors  101  range from about 10 mA to about 1 A at the exemplary frequency of 13.56 MHz. The skin effect, where only the outer part of the conductor  101  contributes to current conduction, typically operates to increase resistance for high frequency currents. Smaller cross-sectional area for conductors  101  results in higher specific resistance thereby increasing the resistance losses in coils  115  and  125 . Typically, a higher resistance for a given inductance lowers the quality factor (Q) of an antenna circuit. Typical values for Q for exemplary embodiments in accordance with the invention are in the range from about 10 to about 40. However, the width of conductors  101  for a given area for planar loop antenna  110  is limited by the requirement that the middle of coils  115  and  125  be conductor free for effective H-field transmission or reception. 
         [0024]    The spacing between conductors  101  of coils  115  and  125  is typically determined by the commercially available process which typically results in a spacing between conductors  101  on the order of about 0.1 mm in an embodiment in accordance with the invention. There is a proximity effect between conductors  101  when carrying an AC current. Each trace of conductor  101  produces an H-field which reduces the useable cross-section of conductors  101  for carrying current and increases the effective resistance. The proximity effect increases with frequency and decreases with increased spacing between conductors  101 . Hence, a closer spacing of conductors  101  increases the resistance of planar loop antenna  110 . 
         [0025]    If an AC current is driven in coil  115 , coil  115  emits an H-field. For illustrative purposes,  FIG. 2   a  shows the H-field for circular loop antenna  215  which can be calculated using the Biot-Savart law. The radial distance r between the center of circular loop antenna  215  and any point in space is given by: 
         [0000]        r ( x,y,z,θ )=√{square root over (( a  cos θ− x ) 2 +( a  sin θ− y ) 2   +z   2 )}{square root over (( a  cos θ− x ) 2 +( a  sin θ− y ) 2   +z   2 )}  (1)
 
         [0000]    where a is the radius of circular loop antenna  215  and θ is the angle between the radius and the x-axis. The z component of the H-field, H z , can be calculated at any point (x,y,z) using the following equation: 
         [0000]    
       
         
           
             
               
                 
                   
                     
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         [0000]    where β is the phase constant 2πf c /c and I A  is the current in the antenna. 
         [0026]    For coils  115  and  125  of planar loop antenna  110  that have a rectangular shape in an embodiment in accordance with the invention, the H-field is typically computed using High Frequency Structural Simulator (HFSS) available from ANSYS Corporation. Typical operating voltages for the contactless reader antenna are typically in the range of about 30 volts to about 40 volts with a current on the order of several 100 mA. 
         [0027]    In a plane parallel and below coil  115 , the magnetic flux in the plane under the center of coil  115  has one direction while the magnetic flux in the plane outside of coil  115  points in the opposite direction (e.g. see direction for H-field of circular loop antenna  215  in  FIG. 2   a ). The flux density is non-homogeneous. Coil  125  is placed relative to coil  115  in such a way (e.g. see antenna overlap  155  in  FIG. 1   b ), that the magnetic flux generated by coils  115  and  125  in one direction is substantially the same as the magnetic flux generated by coils  115  and  125  in the opposite direction so that the magnetic flux substantially cancels and the induced voltage in one coil due to the magnetic field of the other coil is substantially zero. This provides a “zero” coupling antenna in accordance with the invention. 
         [0028]    The coupling coefficient k between coils  115  and  125  may be estimated as follows. A constant AC voltage U 1  is applied to coil  115  having an inductance L 1  and the induced voltage U 2  is measured in coil  125  having an inductance L 2 . Then the coupling coefficient k is given by: 
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         [0000]    The criteria for a “zero” coupling antenna in accordance with the invention is that k≦10%. 
         [0029]    In the active card mode operation of a transceiver device, such as a Near Field Communication (NFC) device, planar loop antenna  110  is connected to the integrated circuit chip comprising the driver circuit (e.g. an NFC chip) such that common ground  150  is connected to connection point  130  between coils  115  and  125 . The driver output of the integrated circuit is connected to common ground  150  and end pad  135  of coil  115  and is used to drive the active load modulation signal. The receiver input of the integrated circuit is connected to common ground  150  and end pad  145  of coil  125  and is used to sense the 13.56 MHz carrier phase of the contactless reader. 
         [0030]      FIG. 2   b  shows induced voltage (Vpp)  224  in coil  125  (see  FIG. 1   b ) as measured between common ground  150  and end pad  145  due to the 13.56 MHz driver output fed into coil  115  as a function of antenna overlap  155  (length of overlap between coils  115  and  125 ) for planar loop antenna  110 . The driver output is connected between common ground  150  and end pad  135  (see  FIG. 1   b ) and applying an alternating current of 60 mA (rms) for the example shown in  FIG. 2   b .  FIG. 2   b  is used to determine the overlap  155  between antenna  115  and  125  that produces the minimum coupling between coils  115  and  125  (i.e. the minimum induced voltage in coil  125 ). Here, planar loop antenna  110  has exemplary dimensions of about 30 mm by about 17 mm with each coil  115  and  125  having dimensions of about 17.5 mm by about 17 mm. Induced voltage  224  in  FIG. 2   b  is shown to have a minimum for antenna overlap  155  being about 5 mm which results in about a 29% overlap in area between coils  115  and  125 . 
         [0031]    To make planar loop antenna  110  insensitive to the influence of metallic objects nearby and thereby reduce unwanted harmonic emissions a layered structure (see  FIGS. 3 and 4 ) is typically used for planar loop antenna  110 . 
         [0032]      FIGS. 3   a - h  and  FIG. 4  in a side view show the layers of an embodiment of planar loop antenna  110  in an embodiment in accordance with the invention. In an embodiment in accordance with the invention, the layers may be connected to each other using an adhesive or, in another embodiment in accordance with the invention, the layers may be laminated together using typical lamination processes used to make smartcards. 
         [0033]      FIG. 3   a  shows top adhesive layer  310  which typically is an adhesive layer made from FASSON S490 adhesive, for example and having a typical thickness of about 10 μm. Top adhesive layer  310  allows planar loop antenna  110  to be easily mounted on the inside of a device such as a smartphone. Alternatively, top adhesive layer  310  may be a foil such as polyethylene terephthalate (PET) with an adhesive such as FASSON S490 being applied to both sides of the foil. Selection of the adhesive material for layer  310  is typically important as the properties of the adhesive should not adversely impact the H-field such as producing absorption of the H-field. 
         [0034]      FIG. 3   b  shows coil antenna  115  having a typical thickness of about 18 μm, typically made from a conductive material such as copper on face  321  of substrate  320 . Substrate layer  320  is typically made from polyethylene terephthalate (PET) foil having a typical thickness of about 38 μm. Alternatively, substrate layer  320  may be made of PVC. In accordance with the invention, it is typically desirable to have the coil antenna  115  and coil antenna  125  lying in parallel planes that have minimal vertical separation from one another.  FIG. 3   c  shows coil antenna  125  which is on opposite face  322  of substrate  320  from face  321 . 
         [0035]    Coil antennas  115  and  125  may be etched antennas, wire antennas, galvano-antennas or printed antennas. For example, for etched antennas, substrate  320  made of PVC having a copper layer (typical thickness of about 1.8 μm) on both sides of substrate  320  may be used. Photoresist material is placed over the copper layers on each side of substrate  320 . A photographic process then projects the antenna coil layout onto the photoresist residing on top of the copper layers on each side of substrate  320 . Using a chemical process, the exposed photoresist is removed, leaving the layout for coils  115  and  125  in the copper layers. A chemical etch then removes the exposed copper leaving only the copper layouts covered by the photoresist material. The photoresist is then chemically removed to yield planar coils  115  and  125 . Coil antennas  115  and  125  may be electrically connected by drilling a hole and filling the hole with conductive paste to create connection  150 . 
         [0036]      FIG. 3   d  shows second adhesive layer  330  having a typical thickness of about 10 μm and typically made from the same material and the same thickness as top adhesive layer  310 .  FIG. 3   e  shows ferrite layer  340  with a typical thickness of about 100 μm and is typically a ferrite foil such as FSF161 (available from MARUWA Co., Ltd. of Japan) which has a real part relative permeability of about 135 and an imaginary part relative permeability less than about 10 at 13.56 MHz . Hence, ferrite layer  340  has a higher magnetic permeability than air and acts to block (magnetic shielding) the H-field from passing through it. This is useful if planar loop antenna  110  is to be positioned over a metal area, such as a battery pack in a smart phone. Without ferrite layer  340 , a metal area proximate to the antenna would typically significantly attenuate the 13.56 MHz alternating H-field. Note that ferrite layer  340  increases the inductance of the antenna equivalent circuit and so has to be taken into account for the antenna matching. More information regarding the effects and design of a ferrite layer, in particular for use in an NFC transponder, may be found in “Design of 13.56 MHz Smartcard Stickers with Ferrite for Payment and Authentication”, Near Field Communication (NFC), 2011 3 rd  International Workshop on, pages 59-64, 2011, which is incorporated herein by reference in its entirety. 
         [0037]      FIG. 3   f  shows third adhesive layer  350  having a thickness of about 10 μm and typically made from the same material as top adhesive layer  310 .  FIG. 3   g  shows second substrate layer  360  having a typical thickness of about 38 μm. 
         [0038]    Finally,  FIG. 3   h  shows metal shield layer  370  having a typical thickness of about 18 μm attached underneath second substrate  360 . Metal shield  370  is typically made from aluminum or copper. Metal shield layer  370  makes planar loop antenna  110  more resistant against de-tuning caused by the presence or absence of various materials behind planar loop antenna  110  as ferrite layer  340  only blocks a portion of the H-field and part of the H-field passes through ferrite layer  340 . The presence or absence of metal (e.g. battery pack) changes the equivalent circuit element values of planar loop antenna  110 . For example, if a fixed matching network is used to match planar loop antenna impedance at a frequency of 13.56 MHz to the integrated circuit amplifier output impedance, the result would be an impedance mismatch. Metal shield layer  370  is already taken into account by the fixed matching network so planar loop antenna  110  is less sensitive to the presence or absence of nearby metal objects. Additionally, metal shield layer  370  provides shielding from electrical fields from other parts of the transponder device or contactless reader at the cost of a reduction in contactless performance. The reduction in contactless performance typically results because the H-field penetrating through ferrite layer  340  produces eddy currents in metal shield layer  370  that generate H-fields that oppose the applied H-field, resulting in an overall reduction of the applied H-field. 
         [0039]    The layer structure of planar loop antenna  110  in accordance with the invention also provides directionality as the H-field emission occurs preferentially in the direction away from metal shield layer  370  as shown in  FIGS. 5   a  and  5   b .  FIG. 5   a  shows the contours of H-field  510  in cross-sectional plane perpendicular to coils  115  and  125 . H-field  510  in  FIG. 5   a  is the magnetic H field for coils  115  and  125  separated by substrate  120  without any additional layers and H-field  510  is symmetrical about substrate  120 . H-field  520  in  FIG. 5   b  is the magnetic H field for coils  115  and  125  using layer structure  450  shown in  FIGS. 4 and 3   a - h.  In contrast to H-field  510  in  FIG. 5   a , H-field  520  in  FIG. 5   b  is asymmetric with H-field  520  being stronger above layer structure  450  and weaker below layer structure  450 . This asymmetry is typically due to the presence of metal shield layer  370  and ferrite layer  340  in layer structure  450  which typically function as magnetic shields. 
         [0040]    While the invention has been described in conjunction with specific embodiments, it is evident to those skilled in the art that many alternatives, modifications, and variations will be apparent in light of the foregoing description. Accordingly, the invention is intended to embrace all other such alternatives, modifications, and variations that fall within the spirit and scope of the appended claims.