Abstract:
The present disclosure relates to a method and an antenna for transmitting/receiving a RF signal at a plurality of different frequencies. Transmitting/receiving a RF signal at a plurality of different frequencies is achieved by providing a F antenna comprising a plurality of switches which can be used to adjust the resonant frequency of the antenna. By providing a F antenna, the antenna will be much smaller than the wavelength at which the antenna is operating. This allows the antenna to be used in compact devices such as PDA&#39;s and cellular phones.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS  
       [0001]    This application claims the benefit of U.S. Provisional Patent Application No. 60/470,025 filed May 12, 2003, the disclosure of which is hereby incorporated herein by reference.  
         [0002]    The present document is related to the co-pending and commonly assigned patent application documents entitled “RF MEMS Switch With Integrated Impedance Matching Structure” U.S. Patent Application No. 60/470,026 filed on May 12, 2003, and “RF MEMS-Tuned Slot Antenna and a Method of Making Same”, U.S. Patent Application No. 60/343,888 filed Dec. 27, 2001 and its related non-provisional application U.S. patent application Ser. No. 10/192,986, which claims priority to U.S. Serial No. 60/343,888. The contents of these related applications are hereby incorporated by reference herein. 
     
    
     
       TECHNICAL FIELD  
         [0003]    The technical field of this disclosure relates to tunable antennas and more specifically, a compact tunable F antenna.  
         BACKGROUND  
         [0004]    Antennas that rely on the opening and closing of switches that are co-located with the antenna for tuning are well known in the prior art. An example of a MEMS tuned slot antenna used for frequency tuning is described in a co-pending U.S. Patent Application (See document number 1 below). The MEMS tuned slot antenna disclosed therein contains a slot that is shorted at one end and open at the other end, with a MEMS switch serving as the short across the open end, to determine the effective length of the slot. By closing different switches along the length of the slot, the frequency of the antenna can be tuned. At resonance, the slot measures one-half wavelength long from the closed end to the first closed MEMS switch. This antenna represents an improvement over previous tunable antenna designs because the current was forced through the switch due to the open end of the slot, thus eliminating any unwanted current paths through the ground plane. However, the effective size of this antenna is dependent on the wavelength, which can create problems when a compact antenna is needed. In general, to make any effective MEMS-tuned antenna, the MEMS switch should provide the only path for one part of the antenna current, because the finite inductance of the switch can be shorted by other nearby metal structures, particularly continuous ground planes.  
           [0005]    Other types of MEMS tuned antennas include patch designs, such as those described in document numbers  7  and  8  (identified below), as well as dipole, and various others. These designs are not preferred because patches, dipoles, and many other antennas are tuned by adding small metal regions that extend the length of the primary metal region. When tuning is performed with MEMS switches, this often causes interference from the DC bias lines. Therefore, it is necessary that the tuning be accomplished by shorting a metal object to a large ground plane, which can serve as both a RF and DC ground. In this way, the DC bias lines can be printed along this ground plane in such a way that they have very high or very low RF impedance, so that they cause minimal interference or coupling to the radiation. The slot antenna discussed above is an ideal candidate, but it suffers from a large size. It also requires that the ground plane be extended on all edges except one, which is left open for tuning.  
           [0006]    Thus, the two important properties for a MEMS-tuned antenna are that the MEMS switch should be the only path for the particular portion of the antenna current that provides the tuning, and the switch should be able to be attached to a large ground plane to avoid interference or coupling from the DC bias. Another important property for many portable electronics or other compact devices is that the antenna should be small compared to the operating wavelength. One antenna that embodies these features is known as an F antenna. It typically consists of a metal wire or strip lying adjacent to the edge of a ground plane, with two connecting posts, one post acting as a feed for the metal strip, and the other acting as a short for impedance matching purposes. Reference 9 below discloses an F antenna by using a loop section for tuning instead of tuning the antenna itself. This design is not nearly as elegant or flexible, as the antenna does not provide a wide and arbitrary tuning range.  
           [0007]    The disclosed antenna addresses the aforementioned needs by providing a simple, compact tunable antenna that is suitable for handheld or portable applications. The antenna can be tuned over a broad frequency range, and the size of the antenna is not solely dependent on the operating wavelength of the antenna such as is the case with typical prior art antennas.  
         DESCRIPTION OF RELATED ART  
         [0008]    1. D. Sievenpiper, “RF MEMS-Tuned Slot Antenna and a Method of Making Same”, U.S. Patent Application Serial No. 60/343,888 and U.S. patent application Ser. No. 10/192,986, which is related to 60/343,888. These applications describe a tunable slot antenna. The presently disclosed technology is different in that the presently disclosed technology allows an antenna to be much smaller than the operating wavelength which can be important for certain handheld and/or portable applications.  
           [0009]    2. 1. Korisch, “Planar Dual Frequency Band Antenna”, U.S. Pat. No. 5,926,139 describes a basic planar RF antenna and includes meander line type structures for setting the resonant frequency.  
           [0010]    3. S. Moren, C. Rowell, “Trap Microstrip PIFA”, U.S. Pat. No. 6,380,895. This patent describes another type of planar RF antenna, and also includes meander line structures for setting the resonant frequency.  
           [0011]    4. N. Johansson, “Antenna Device and Method for Portable Radio Equipment”, U.S. Pat. No. 6,016,125. This patent describes an antenna that is tunable or reconfigurable by adjusting the position of a whip portion, which contacts an impedance matching inductor. This could be used either to adjust the position of the antenna to improve the impedance match, or presumably to tune the resonant frequency of the antenna. However, this antenna requires physical control of the antenna position by a user, and the antenna is largely stationary.  
           [0012]    5. Y. J. Chen, H. J. Li, R. B. Wu, “Multi-Resonance Horizontal U-Shaped Antenna”, U.S. Pat. No. 5,644,319. This patent describes a multi-resonant antenna, however the antenna is not tunable. Furthermore, the antenna requires a folded structure that increases the size of the antenna.  
           [0013]    6. Hiroshi Okabe, Ken Take, “Tunable Slot Antenna with Capacitively Coupled Island Conductor for Precise Impedance Adjustment”, U.S. Pat. No. 6,034,655. This patent describes a slot antenna using a cavity structure. The cavity structure increases the size of the antenna significantly, and the use of a closed-end slot forbids the use of MEMS switches.  
           [0014]    7.Robert Snyder, James Lilly, Andrew Humen, “Tunable Microstrip Patch Antenna and Control System Therefore”, U.S. Pat. No. 5,943,016 describes a method of using a patch antenna by using RF switches to connect or disconnect a series of tuning stubs. However, this antenna is extremely sensitive to the position of the bias circuits and does not have the ability to tune the polarization and the pattern.  
           [0015]    8. Jeffrey Herd, Marat Davidovitz, Hans Steyskal, “Reconfigurable Microstrip Array Geometry which Utilizes Microelectromechanical System MEMS switches”, U.S. Pat. No. 6,198,438 describes an array of patch antennas that are connected by RF MEMS switches. This antenna can be selectively tuned by turning on or off various switches to connect the patches together. Larger or smaller clusters of patches will create antennas operating at lower or higher frequencies. However, this antenna requires a large number of switches and the antenna does not provide a way to eliminate the problem of interference between the DC feed lines and the RF part of the antenna.  
           [0016]    9. Gerard Hayes, Robert Sadler, “Convertible Loop/Inverted F Antennas and Wireless Communicators Incorporating the Same”, U.S. Pat. No. 6,204,819 describes an F-type antenna. However, this antenna has significant drawbacks due to its complexity. The antenna requires each separate frequency of operation to be addressed by a different type of antenna (loop, F, etch). This requires a different set of design equations for different resonant frequencies and modes of operation. Furthermore, this antenna does not allow for angle diversity.  
           [0017]    10. De Los Santos “Tunable Microwave Network Using Microelectromechanical. Switches” U.S. Pat. No. 5,808,527 describes a MEMS switch for tuning, but does not discuss integration of a switch into an antenna.  
           [0018]    11. Lam, Tangonan, and Abrams, “Smart Antenna System Using Microelectromechanically Tunable Dipole Antennas and Photonic Bandgap Materials” U.S. Pat. No. 5,541,614 describes an antenna system using microelectromechanically tunable dipole antennas and photonic bandgap materials.  
         SUMMARY  
         [0019]    The presently disclosed technology provides an F type antenna that addresses the aforementioned needs. The antenna is much more compact than previous designs and has the ability to match the input impedance to a 50 ohm transmission line over a broad tuning bandwidth. This is primarily due to the simple resonant structure that provides the mode or modes of radiation. The tuning mechanism of the present invention is also compatible with MEMS switch devices. Previous switches were somewhat lossy, which results in a low-efficiency antenna. This effect is aggravated by high-Q antennas, and thus rules out tunable F-type antennas, which are typically high Q. The compact nature of the F-type antenna could allow it to be used in, for example, a handheld transceiver or for in-car communications with a PDA or telephone. Also, the ability to tune the resonant frequency would allow a single antenna to be installed in cars that are sold in different countries, since the antenna could simply be tuned to use the frequencies allocated for each service in each individual country. Other services that could benefit from such an antenna are AMPS, PCS, Bluetooth, 802.1 1a, or military bands.  
           [0020]    An embodiment of a tunable F antenna for transmitting/receiving a RF signal at a desired one of a plurality of different frequencies is disclosed. The antenna comprises an electrically conductive tab positioned along a conductive sheet. A plurality of switches is provided which act when closed to couple the conductive sheet to the electrically conductive tab. The plurality of switches are closable in a controlled manner to change a desired resonant frequency at which the antenna transmits/receives the RF signal. A feed line coupled to the electrically conductive tab is provided for coupling the RF signal to/from the electrically conductive tab.  
           [0021]    Other embodiments of a tunable F antenna for transmitting/receiving a RF signal at a desired one of a plurality of different frequencies are disclosed. The antenna comprises an electrically conductive tab positioned along a conductive sheet. A plurality of switches is provided which act when closed to couple the conductive sheet to the electrically conductive tab. The plurality of switches are closable in a controlled manner to change a desired resonant frequency at which the antenna transmits/receives the RF signal. The plurality of switches is also positioned so as to allow adjustment of the radiation pattern of RF signal. A feed line coupled to the electrically conductive tab is provided for coupling the RF signal to/from the electrically conductive tab. 
       
    
    
     BRIEF DESCRIPTIONS OF THE FIGURES  
       [0022]    [0022]FIG. 1 a  shows the front side of an antenna according to one embodiment of the present invention.  
         [0023]    [0023]FIG. 1 b  shows the backside of the antenna depicted in FIG. 1 a .    
         [0024]    [0024]FIG. 1 c  shows an embodiment of the antenna of FIG. 1 a  sized to be received inside a handheld device.  
         [0025]    [0025]FIG. 2 a  shows a transparent view of a switch which may be used in the present invention.  
         [0026]    [0026]FIG. 2 b  shows a transparent view of a switch which may be used in the present invention.  
         [0027]    [0027]FIG. 3 a  shows a simplified diagram of the antenna depicted in FIG. 1 a.    
         [0028]    [0028]FIG. 3 b  shows the relationships between the components of the equivalent circuit of FIG. 3 c  and the model of FIG. 3 a.    
         [0029]    [0029]FIG. 3 c  shows the equivalent circuit for the antenna depicted in FIG. 3 a.    
         [0030]    [0030]FIGS. 4 a - 1  through  4   f - 2  show the simulated and measured resonant frequencies for the antenna depicted in FIG. 3 a  for different switch positions.  
         [0031]    [0031]FIGS. 5 a  and  5   b  show an alternate embodiment for placing the electrically conductive tab relative to the conductive sheet/ground plane.  
         [0032]    [0032]FIG. 5 c  shows how the switch is coupled to the electrically conductive tab and the conductive sheet/ground plane when using the embodiment depicted in FIG. 5 b.    
         [0033]    [0033]FIG. 5 d  shows an embodiment of providing an electrically conductive tab having different thicknesses between switches.  
         [0034]    [0034]FIG. 6 shows an alternate embodiment for the electrically conductive tab.  
         [0035]    [0035]FIG. 7 a  shows a graph of the resonant frequencies of the antenna for each side of the antenna for different switch positions.  
         [0036]    [0036]FIG. 7 b  shows where the antenna depicted in FIG. 1 a  emits the two modes.  
         [0037]    [0037]FIG. 7 c  shows how the radiation pattern can be changed depending on which switches are closed. 
     
    
     DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS  
       [0038]    This technology will now be described more fully hereinafter with reference to the accompanying drawings, in which preferred embodiments are shown. The presently described technology may be embodied in many different forms and should not be construed as limited to the embodiments set forth herein. Further, the dimensions of certain elements shown in the accompanying drawings may be exaggerated to more clearly show details. The present disclosure should not be construed as being limited to the dimensional relations shown in the drawings, nor should the individual elements shown in the drawings be construed to be limited to the dimensions shown.  
         [0039]    [0039]FIG. 1 a  depicts a front side view of an F antenna according to the present disclosure. The antenna, in its most basic form, comprises an electrically conductive tab  2 , a conductive sheet or ground plane  4 , a feed line  6 , and switches  8 . F antennas can be broadly characterized as typically having an antenna size between ¼-½ the wavelength of the operating frequency of the antenna. Due to the small size of F antennas, the components may be conveniently mounted on dielectric substrate  12  preferably provided by a circuit board such as those used in small electronic devices, such as a portable handset device, cellular telephone, PDA, or other communication device  20 , as shown by FIG. 1 c . However, those skilled in the art will realize that the antenna according to the presently disclosed technology can be integrated into a variety of devices and is not limited to portable handset devices. The components of the antenna will now be described in more detail.  
         [0040]    Since the antenna of FIG. 1 a  can be used in portable handheld devices, it is to be appreciated that the antenna of FIG. 1 a  may be sized for use in such applications. FIG. 1 c  shows an embodiment of the antenna of FIG. 1 a  sized for use in a handheld device  20 .  
         [0041]    The antenna comprises an electrically conductive tab  2 , preferably formed by etching a metal, such as copper, conventionally used on commercially available circuit boards  12 . The conductive sheet  4  can also be conveniently etched from the same metal. The electrically conductive tab  2  can be used to transmit or receive a RF signal. If the electrically conductive tab  2  is used to transmit a RF signal, it will receive the RF signal to be transmitted from the feed line  6  (preferably implements by a microstrip line) mounted on the backside of the printed circuit board  12 . The feed line  6  is shown as a dashed line in FIG. 1 a , to indicate its position relative to the electrically conductive tab  2 , conductive sheet  4 , and switches  8 . In order to transmit a RF signal, one of the switches  8  (discussed later) should electrically short the electrically conductive tab  2  and the conductive sheet  4 . Also, the positioning of the switch  8  should provide a resonance which is substantially the same as the RF signal to be transmitted. This will be discussed in further detail later.  
         [0042]    Similarly, if the antenna is used to receive a RF signal, the position of the switches  8  should provide a resonance with corresponds to the RF signal to be received. When a RF signal is received, the electrically conductive tab  2  couples the received RF signal into the feed line  6 , where it can be coupled into other components for further processing. Shown in FIG. 1 a  are three switches  8 , however, the actual number of switches used is a design consideration as will be discussed later. Furthermore, it will become apparent that by providing multiple switches at different locations along the conductive metal tab  2 , the antenna may be tuned to transmit or receive multiple RF signals.  
         [0043]    [0043]FIG. 1 b is a rear view of the antenna of FIG. 1 a , depicting the feed line  6  and switch actuating lines  10  on the backside of the circuit board  12 , together with other circuits  22  that may be used with the antenna. The switch actuating lines  10  are used to activate the switches  8 , as is discussed later. The electrically conductive tab  2 , conductive sheet  4 , and switches  8  are shown in dashed lines to indicate their position on the front side of circuit board  12  relative to the feed line  6  and switch actuating lines  10 . The feed line  6  is connected to the electrically conductive tab  2  through a metal via (not shown) in the circuit board  12 . The feed line  6  can be coupled to the electrically conductive tab  2  at a fixed location anywhere along the longitudinal axis of the electrically conductive tab  2 . Although the electrically conductive tab  2  does not have preferred dimensions, the frequency and passband of the antenna are dependent on its physical dimensions, such as its width and length.  
         [0044]    Located adjacent to the electrically conductive tab  2  is a conductive sheet  4 , as illustrated in FIG. 1 a.  The conductive sheet  4  and electrically conductive tab  2  are connected with switches  8 . To help reduce the size of the antenna, the switches  8  are preferably in the gap between the electrically conductive tab  2  and conductive sheet  4  to eliminate the need for wire bonds or similar structures to link the switches  8  to the electrically conductive tab  2  and conductive sheet  4 . This distance D between the electrically conductive tab  2  and conductive sheet  4  is typically about 1 mm. There is a slight dependence of the bandwidth of the antenna on the distance D; increasing D will increase the bandwidth, but this effect is usually so small as to be immeasurable. Theoretically, D could be increased to provide significantly large bandwidths, however this would put severe constraints on being able to reduce the size of the antenna.  
         [0045]    When one of the switches  8  is activated a short between the electrically conductive tab  2  and the conductive sheet  4  is created. An example of a switch  8  that may be used in this application is described in U.S. Patent Application No. 60/470,026 filed May 12, 2003 mentioned above The switch  8  may be placed on either side of the feed line  6 . The number of switches  8  used is a matter of design and will be discussed later. Because high currents typically pass through the closed switch  8 , the antenna will have high efficiency if the switch  8  has low RF loss. As such, the switch  8  is preferably a RF MEMS switch fabricated on a GaAs substrate using micromachining techniques.  
         [0046]    A close-up views of an exemplary switch  8  are shown in FIGS. 2 a  and  2   b . The portions shown in these views roughly corresponds to the region bounded by dashed line  3  in FIG. 1 a . Only the switch ports and terminals are shown and not the internal switch construction of switch  8  for ease of illustration. The switch  8  preferably has a rectangular layout and includes first and second DC bias ports  14   a ,  14   b , and first and second RF terminals  16   a ,  16   b . The first DC bias port  14   a  is connected through the circuit board  12  in the gap between the electrically conductive tab  2  and conductive sheet  4  its associated control line  6  on the backside of the printed circuit board  12 . The second DC bias port  14   b  is connected to the conductive sheet  4 . The first RF terminal  16   a  is mounted on (and connected to) the electrically conductive tab  2  and the second RF terminal  16   b  is mounted on the conductive sheet  4 . To accommodate this arrangement, the electrically conductive tab  2  may be fabricated with a recess  5  to accommodate the first DC bias port  14   a  as shown in FIG. 2 a , or a protrusion  7  to connect to the first RF terminal  16   a  as shown in FIG. 2 b . The switch  8  is preferably a MEMS type switch of the type that is operated by moving a cantilever beam (not shown), which beam bends downwards to couple the first and second RF terminals  16   a ,  16   b  together when the switch actuating lines  10  provides an actuating voltage between the DC bias ports  14   a ,  14   b . The second DC bias port  14   b  can serve as both a DC and RF ground by connecting the second DC bias port  14   b  to the second RF terminal  16   b  with, for example, wire bonds. In some embodiments, the switch  8  may have as few as three terminals/ports (a ground, a DC bias port and a RF terminal). Like the feed line  6 , the actuating lines  10  are preferably disposed on the backside of the circuit board  12  (See FIG. 1 b ) and are preferably connected to the switches  8  using metal vias  9  through the circuit board  12   
         [0047]    If desired, the switches  8  may be disposed on the backside of the circuit board  12 , in which case the switch actuation lines  10  may connect directly to the first DC bias port  14   a . In that case, metal vias will be preferably used to connect the first and second RF terminals  16   a ,  16   b  to the electrically conductive tab  2  and conductive sheet  4 , respectively, and connect the second DC bias port  14   b  to the conductive sheet  4 . In either case, the switch  8  is preferably sealed in a package and may be electrically connected to the circuit board  12  using a variety of well-known techniques such as flip chip bonding, wave soldering, or wire bonding.  
         [0048]    Shown in FIG. 3 a  is a simplified diagram of the antenna depicted in FIGS. 1 a  and  1   b . This simplification is for modeling purposes only, but the concepts described below are applicable to the larger conductive sheet  4  depicted in FIGS. 1 a  and  1   b . The complete equivalent circuit for the simplified antenna is depicted in FIG. 3 c  and the relationships between the equivalent circuit of FIG. 3 c  and the model of FIG. 3 a  is depicted by FIG. 3 b . In the simplified diagram of FIG. 3 a , the antenna is assumed to comprise a symmetric pair of metal strips, functioning as an electrically conductive tab  2  and a conductive sheet  4 . In the antenna shown in FIG. 3 a , the total width (W) of the electrically conductive tab  2  and conductive sheet  4  is normalized to one. The width (W) of the electrically conductive tab  2  effectively determines the size of the antenna. A feed line  6  is coupled to the electrically conductive tab  2  and a closed switch  8  is used to create a connection between the feed line  6  and conductive sheet  4 . Typically, for a given antenna, the feed line  6  is located at a fixed position, so the antenna parameters will depend on the position of the closed switch  8  relative to the position of the feed line  6 . One important difference between this antenna and the previously discussed slot antennas is the fact that the size of this antenna can be made much smaller than the operating wavelength. This has significant advantages for portable devices and other applications where compact antennas are required. For example, when the electrically conductive tab  2  has a width between 5-6 cm, the antenna has been shown to resonate at 900 MHz, 1.9 GHz, and 2.45 GHz. An antenna size (width of the conductive metal tab  2 ) of 5-6 cm operating at 2.45 GHz may be comparable to current state of the art devices, however, current state of the art devices operating at 900 MHz require an antenna size on the order of 15 cm. In addition, by varying the capacitive and inductive properties of the antenna using the techniques described herein, higher and lower resonant frequencies can be produced using the same electrically conductive tab  2 . As a result, it is clear that the size of the antenna described herein can be fixed and made independent of the RF signal being transmitted or received with a given frequency range. Thus, the size of the antenna can remain small. This is a result of the fact that the present antenna relies on embedded resonant structures that can be modeled as the lumped circuit elements shown in FIG. 3 b  and discussed below.  
         [0049]    The portion of the electrically conductive tab  2  and conductive sheet  4  located to the left (L) of the feed line  6  can be modeled by inductor L 1 , and the portion of the electrically conductive tab  2  and conductive sheet  4  located to the right (R) of the switch  8  when closed can be modeled by inductor L 2 . The region between electrically conductive tab  2  and conductive sheet  4 , to the left of the feed line  6 , and to the right of the closed switch  8 , can be modeled as capacitors C 1  and C 2 , respectively. Finally, the region between the electrically conductive tab  2  and conductive sheet  4 , and between the feed line  6  and closed switch  8 , can be modeled as inductor L 3 , while the capacitance of that region is neglected. Resistors R1 and R2 act as radiation dampers. Vs is the signal the feed line  6  provides to the electrically conductive tab  2 . The presence of L 1 , C 1 , and L 2 , C 2  produce two main resonant frequencies. The values of L 1 , L 2 , L 3 , C 1 , C 2 , R 1 , and R 2  can then be used to predict the behavior of the antenna, specifically the resonant frequencies of the antenna.  
         [0050]    The values of L 1 , L 2 , L 3 , C 1 , C 2 , R 1 , and R 2  can be approximated by determining the capacitance/unit length (Eq. 1) and inductance/unit length (Eq. 2).  
               Capacitance        /        unit                 length     =       width                   (       eps                 1     +     eps                 2       )         π   *   Arc                   Cosh        (     a   /   g     )                   Eq   .              1                               
 
         Inductance/unit length=Capacitance/unit length*(Characteristic Impedance) 2    Eq. 2  
         [0051]    Where:  
         [0052]    Characteristic Impedance=377 Ω 
         [0053]    width=Horizontal Width of electrically conductive tab (W)  
         [0054]    eps0=permittivity of free space  
         [0055]    eps1=dielectric constants of the material above antenna (typically air)  
         [0056]    eps2=dielectric constants of the material below antenna (typically the substrate on which the antenna is mounted, i.e. the circuit board)  
         [0057]    a=length of the electrically conductive tab or conductive sheet/ground plane (the (the tab an sheet are both assumed to be symmetric)  
         [0058]    D=size of the gap  
           L   1 =Min[feed line, switch]*Inductance/unit length  
           L   2 =(1−Max[feed line, switch])*Inductance/unit length  
           L   3 =Absolute Value of (feed line−switch)*Inductance/unit length  
           C   1 =Min[feed line, switch]*Capacitance/unit length  
           C   2 =(1−Max[feed line, switch])*Capacitance/unit length  
         [0059]    Min[feed line, switch] is the distance between the feed line  6  or the switch  8 , whichever is smaller with respect to the left most side of the electrically conductive tab  2 , as shown in FIG. 3 a.    
         [0060]    Max[feed line, switch] is the distance between the feed line  6  or the switch  8 , whichever is greater with respect to the left most side of the electrically conductive tab, as shown in FIG. 3 a.    
         [0061]    Since the resonant frequencies of the antenna are determined by the Capacitance/unit length and the Inductance/unit length, one can design an antenna for any frequencies of interest by varying these parameters. Furthermore, the total impedance (z) of the antenna can be calculated using Equation 3.  
             z   =     1         1   /   z                   1     +       1   /   z                   2     +       1   /   z                   3                 Eq   .              3                               
 
         [0062]    where  
           z                 1     =       j                 ω                 L                 1     +     1     j                 ω                 C1       +   R       ;             z2   =       j                 ω                 L2     +     1     j                 ω                 C2       +   R       ;              and           z3   =     j                 ω                   L3   .                             
 z3=jωL3.  
         [0063]    R, which is the same as R 1  and R 2  shown in FIG. 3 c , is the radiation resistance, which is somewhat arbitrary. The behavior of the antenna is determined primarily by the frequencies of two main resonances, and R mainly determines the bandwidth of these different resonances. It typically has a value of more than a few ohms, but much less than 377 ohms. The value of co is the angular frequency of the signal provided by the feed line  6 .  
         [0064]    Finally, using the values of z, the magnitude of the reflection for various switch positions can be determined by using equation 4. Equation 4 is the formula for the reflection in a 50-ohm transmission line that is terminated by impedance, z.  
         Reflection=20*log [ Abs [(50 −z )/(50 +z )]]  Eq. 4  
         [0065]    Shown in FIGS. 4 a - 1  through  4   f - 2  are simulated graphs of the expected resonant frequencies as well as the measured resonant frequencies for various switch positions using the antenna depicted in FIG. 3 a . Initially, the feed line  6  is fixed at a distance ¼ L away from the left edge with the following parameters.  
         [0066]    Characteristic Impedance=377 Ω 
         [0067]    width (W)=7.5 cm  
         [0068]    eps0=8.85×10 −12    
         [0069]    eps1=eps0  
         [0070]    eps2=4×eps0  
         [0071]    a=1 cm  
         [0072]    D=1 mm  
         [0073]    R=20 Ω 
         [0074]    In the graphs depicted in FIGS. 4 a - 1  through  4   f - 2 , the x-axis represents the frequencies, and the y-axis represents the reflection (return loss). As will be seen, the return loss is significantly lower at the resonant frequencies. Also, as the position of the switch  8  moves from the left side of the antenna towards the right side. We can observe changes in the frequencies of the two main modes, which are associated with the capacitors C 1 , C 2 , combined with inductors L 1 , L 2 , L 3 , which radiate energy into free space as modeled by radiation resistors R 1  and R 2 . When the switch  8  is near the left edge, the resonant frequency associated with C 1  and L 1  is high, while the resonant frequency associated with C 2  and L 2  is low. This is because of the relatively larger capacitance and inductance associated with C 2  and L 2  when the switch  8  is near the left edge.  
         [0075]    [0075]FIG. 4 a - 1  is the simulated results and FIG. 4 a - 2  depicts the measured results for an embodiment where the switch  8  is located at a distance {fraction (1/16)} W away from the left edge and a single resonant frequency associated with C 2  and L 2  is seen near 1 GHz. The resonant frequency associated with C 1  and L 1  is too high and cannot be seen in FIGS. 4 a - 1  and  4   a - 2 . As the switch  8  is moved toward the feed line  6 , the resonance associated with C 1  and L 1  shifts lower because the change in placement of the switch  8  causes the values of C 1  and L 1  to increase. FIG. 4 b - 1  is the simulated results and FIG. 4 b - 2  depicts the measured results for an embodiment where switch  8  is located at a distance {fraction (3/16)} W away from the left edge of the antenna. The resonance previously seen around 1 GHz has moved up in frequency slightly, and a second resonant frequency associated with C 1  and L 1  is seen near 4 GHz.  
         [0076]    [0076]FIG. 4 c - 1  is the simulated results and FIG. 4 c - 2  depicts the measured results for an embodiment where the switch  8  is located a distance {fraction (5/16)} W away from the left side. As can be seen, the two resonant frequencies broaden and move closer to each other, because the switch has moved past the feed line  6 . As the switch  8  moves past the feed line  6  the two resonant frequencies continue moving towards each other (See FIG. 4 d - 1  which depicts the simulated results and FIG. 4 d - 2  which depicts the measured) until the switch  8  is symmetric to the feed line  6  (i.e. located a distance ¾ W away from the left edge). At this point the two resonant frequencies merge into a single resonance as shown in FIGS. 4 e - 1  (depicting the simulated results) and 4 e - 2  (depicting measured results). Then, as the switch  8  moves closer to the right edge, the two resonant frequencies cross, as shown in FIGS. 4 e - 1  (depicting the simulated results) and 4 f - 2  (depicting measured results), where the switch  8  is located a distance {fraction (13/16)} W away from the left edge. Now the resonance associated with C 2  and L 2  is higher in frequency because the values for C 2  and L 2  decrease as the switch  8  moves closer to the right side of the antenna  1 . As shown in FIGS. 4 f - 1  and  4   f - 2 , the resonance associated with C 2  and L 2  is approximately 6 GHz, while the resonance associated with C 1  and L 1  is around 3.5 GHz. In this way it can be seen that a plurality of switches  8  may be provided at various positions along the conductive metal tab  2  to provide a plurality of resonances.  
         [0077]    Since the values for C 1 , C 2 , L 1 , and L 2  partially determine the resonances associated with the antenna, one can design an antenna of this type for any resonances by varying the values for Capacitance/unit length and Inductance/unit length. One way of lowering the Capacitance/unit length to increase the bandwidth of the resonant frequencies, is to place the electrically conductive tab  2  further away from the conductive sheet  4  as shown in FIG. 5 a . In this case, fingers  18  are extended from the electrically conductive tab  2  to the switches  8 . Of course, it would also be possible to extend fingers from the conductive sheet  4  up to the switches  8 . If the fingers  18  are made sufficiently narrow they will not significantly add to the capacitance. In addition, the distance between the electrically conductive tab  2  and conductive sheet  4  can be different in the regions between the switches  8  as shown in FIG. 5 d.    
         [0078]    In order to increase the Capacitance/unit length so as to lower the resonant frequencies for a given width of the electrically conductive tab  2 , the electrically conductive tab  2  and conductive sheet  4  can be made to overlap on opposite sides of the circuit board as shown in FIG. 5 b . A recessed area is made in either the electrically conductive tab  2  or conductive sheet  4  (shown in the conductive sheet  4  in FIG. 5 b ) to prevent the electrically conductive tab  2  and conductive sheet  4  from being shorted together. The first and second DC ports  14   a ,  14   b , and the first and second RF terminals  16   a ,  16   b  can be appropriately connected to the electrically conductive tab  2  and conductive sheet  4  either directly, or through metal vias as shown in FIG. 5 c.    
         [0079]    Also, the Inductance/unit length can be increased to lower the resonant frequencies without significantly reducing their bandwidth for a given antenna size, or to increase the magnetic component of the stored field to improve efficiency. Increasing the Inductance/unit length can be accomplished by meandering the electrically conductive tab  2  as shown in FIG. 6 between neighboring switches  8 . Those skilled in the art will realize that both the inductance and capacitance modification structures discussed above can have different geometries in different regions to achieve greater control of the frequency and bandwidth of each resonance.  
         [0080]    If appreciable size is allowed for the width of the electrically conductive tab  2 , such as somewhere between one-quarter and one-half the wavelength of the operating frequency, then the antenna can also be made to have an adjustable radiation pattern. As previously discussed, different resonant modes are associated with different regions in the antenna (e.g. C 1 , L 1 , and C 2 , L 2 ). If these modes are close together, and the antenna is excited at a fixed frequency, then the relative frequencies of the modes can be considered as a phase difference between these various regions in the antenna. An illustrative example of this is further discussed below. If the right side of the antenna (C 2  and L 2 ) leads the left side (C 1  and L 1 ) in phase, then the sum of these modes will result in a beam that is directed to the left. If the right side lags the left, then the beam will be directed toward the right. If they are exactly in phase, then the beam will be directed to the broadside. In each case, the radiation pattern can be further modified by controlling the dielectric constant on either side of the antenna, since the radiation will tend to be stronger on the side with the higher dielectric constant.  
         [0081]    [0081]FIG. 7 a  shows a plot of the resonance frequencies of the two main modes (x-axis) of the antenna as a function of position of the switch  8  (y-axis) for the antenna depicted in FIG. 3 a . The resonance frequencies are labeled as Left Side and Right Side. The resonance designated Left Side is the resonance associated with the left side of the antenna, (i.e. L 1 , C 1 ). The resonance designated Right Side is the resonance associated with the right side of the antenna, (i.e. L 2 , C 2 ). Also shown in FIG. 7 a  are three vertical lines, designated A, B, and C. These lines correspond to switches A, B, C shown in FIG. 7 b . FIG. 7 a  shows the resonant frequencies of the two main modes for the left side and right side when either switch A, B, or C is closed. Switch B is nearly symmetrical with the feed line  6 , and at that point, the two modes cross in frequency. Switches A and C can be placed at several locations near this point, typically within 2-5 mm and used to adjust the radiation pattern. However, those skilled in the art will realize that the actual placement of switches A and C will also depend on the geometry of the antenna and the bandwidth. Depending on which switch  8  is closed, the relative phases of the two main modes, labeled as Mode # 1  and Mode # 2  in FIG. 7 b , can be adjusted, thus changing the radiation pattern. If switch B is closed, then the radiation will be strongest towards the broadside. If switch A or C is closed, then the radiation will be stronger either to the left, or right side, respectively. This concept is illustrated in FIG. 7 c  as three separate beams, and shows how this technique can be used for angle diversity in a multipath environment.  
         [0082]    From the foregoing description, it will be apparent that the presently described technology has a number of advantages, some of which have been described herein, and others of which are inherent in the disclosed embodiments. Also, it will be understood that modifications can be made to the apparatus and method described herein without departing from the teachings of subject matter described herein. For example, the edges of the conductive tab  2  and the conductive sheet  4  in the disclosed embodiment are depicted as being defined by straight lines. However, when installed the disclosed antenna in a handheld device such as a cellular telephone or a personal digital assistant (and in any other communications device), it may prove convenient in such applications to round the corners (or other portions) of the tab  2  and/or the sheet  4 , in order to more easily accommodate the disclosed antenna in a communications device. As such, the tab  2  and sheet  4  do not necessarily need to be limited to the rectilinear embodiments depicted by the figures. For such reasons and others, the disclosed technology is not to be limited to the described embodiments except as required by the appended claims.