Abstract:
A method and apparatus is directed to generating an improved temperature coefficient for the current limit in a switching regulator/driver circuit. The current limit sense circuit includes a comparator that compares two signals to determine when the current limit has been exceeded. One signal is produced from a temperature independent voltage source, a trans-conductance cell, and a sensor resistor circuit. Another signal is produced by an active output circuit, such that the signal corresponds to the current associated with the switching regulator/driver circuit. The current sensed by the regulator/driver is temperature dependent due to the resistances in the active output circuit, the sensor resistor circuit, and the trans-conductance cell. Each of these resistances has a temperature coefficient. The temperature coefficients determine the amount of temperature dependence in the sensed switching/regulator current. The resistance materials are chosen such that the temperature dependence of the sensed current is minimized.

Description:
FIELD OF THE INVENTION 
     The present invention relates to switching regulators and driver circuits. In particular, the present invention relates to a method and apparatus that provides for an improved temperature coefficient for the current limit in a switching regulator circuit. The improved temperature coefficient in the current limit may also be employed for use in a driver circuit such as an RS-232 driver. 
     BACKGROUND OF THE INVENTION 
     Current limit is an important parameter for switching regulator and driver circuits. A precisely controlled value for the current limit in a wide temperature range is always desired. An example of a current sensor circuit ( 100 ) that may be used in a switching regulator circuit is shown in FIG.  1 . Switching regulator circuit  100  includes a comparator ( 110 ), a differential gain amplifier ( 120 ), a reference voltage (V REF ), a bi-polar junction transistor (BJT) (Q 1 ), a diode (D 1 ), an internal voltage reference (V REF(I) ), a sense resistor (R S ), and two resistors (R 1  and R 2 ). 
     Comparator  110  includes a sensor input (SNS) that is coupled to node N 10 , a reference input (REF) that is coupled to node N 11 , and an output that is coupled to node N 16 . Differential gain amplifier  120  includes differential inputs that are coupled to node N 12  and power supply node N PS10 , and an output that is coupled to node N 10 . Internal voltage reference V REF(I)  is coupled between node N 11  and node N 13 . Diode D 1  is coupled between node N 13  and node N 14 . Transistor Q 1  includes a base that is coupled to node N 15 , an emitter that is coupled to node N 14 , and a collector that is coupled to power supply node N PS11 . Resistor R 1  is coupled between node N 15  and power supply node N PS12 . Resistor R 2  is coupled between node N 15  and power supply node N PS11 . 
     The resistors (R 1 , R 2 ) operate as a voltage divider. In this example, sense resistor R S  is located “off-chip” as can be resistors R 1  and R 2 . A circuit ground potential (GND) is coupled to power supply node N PS11 . 
     In operation, comparator  110  produces an output signal when the voltage signal level at node N 10  exceeds the signal level at node N 11 . Differential gain amplifier  120  produces the output signal based on the voltage drop across the sense resistor (R S ). The voltage drop across the sense resistor (R S ) is equal to the product of the value of the sense current (I S ) and the value of the sense resistor (R S ). The differential gain amplifier ( 120 ) then takes the resulting voltage value (I S ·R S ) and scales it (e.g., ×3). This scaled value is outputted by differential gain amplifier  120  at node N 10 . The sensor input (SNS) receives a signal from the output of the differential gain amplifier ( 120 ). 
     Similarly, the reference input (REF) receives a signal from the output of transistor Q 1 . Transistor Q 1  produces the signal based on the voltage present across resistor (R 2 ) of the voltage divider (R 1 , R 2 ) due to the reference voltage (V REF ). Diode D 1  provides an offset to compensate for the voltage drop (V BE ) across the base-emitter junction. The internal reference (V REF(2) ) voltage is provided as a design adjustment. Therefore, the signal received at the reference input (REF) of the comparator ( 110 ) is proportional to the reference voltage (V REF(1) ). The equation for the switching regulator circuit ( 100 ) of FIG. 1, when the sensor current (I S ) is at its peak value, can be expressed as follows: 
     
       
         3· I   S   ·R   S =[( R   2   ·V   REF(1) )/( R   1   +R   2 )]− V   REF(2)   
       
     
     
       
           I   S ={[( R   2   ·V   REF(1) )/( R   1   +R   2 )]− V   REF(2) }/(3· R   S ) 
       
     
     The value of I S  depends on the ratio of R 2 /(R 1 +R 2 ), the value of V REF(1)  and V REF(2) , and R S . Assuming V REF(1)  and V REF(2)  have no appreciable temperature coefficient, the temperature coefficient of I S  is only dependent upon the temperature coefficient of R S . R S  may be an equivalent resistance such as the on resistance (R DS(ON) ) of a MOSFET transistor in a switching regulator. In this instance, the I S ·R S  has a large temperature coefficient that is on the order of 4000 ppm/° C., which is intolerable in some applications. 
     SUMMARY OF THE INVENTION 
     The present invention is directed to a method and an apparatus that improves the temperature coefficient for the current limit in a switching regulator, and also in driver circuits. An improved switching regulator/driver circuit includes “on-chip” resistance circuits that allow for a reduced temperature coefficient associated with the current limit. The improved temperature coefficient of the current limit may be arranged to provide for a constant current limit in the switching regulator or driver circuit. High output currents are limited over a wide range of temperature changes, providing for improved protection to the switching regulator or driver circuit. 
     Briefly stated, a method and apparatus is provided that is directed to generating an improved temperature coefficient for the current limit in a switching regulator circuit. A current limit sense circuit is employed that includes a comparator that compares two signals to determine when the current limit has been exceeded. One signal is produced from an input voltage source that has no temperature coefficient, a trans-conductance cell, and a sensor resistor circuit. An active output circuit produces another signal that corresponds to the current associated with the switching regulator circuit. The current sensed by the regulator is temperature dependent due to the resistances in the active output circuit, the sensor resistor circuit, and the trans-conductance cell. Each of these resistances has a temperature coefficient. The sum of the temperature coefficients determines the amount of temperature dependence in the sensed switching regulator circuit current. The resistance materials are chosen such that the temperature dependence of the sensed current is minimized. Also, all resistors are integrated into a single chip, reducing the costs associated with external pins and external resistor components. A similar arrangement may be applied to a driver circuit with a limited current output. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic diagram illustrating a sensor circuit in a conventional switching regulator circuit. 
     FIG. 2 is a schematic diagram illustrating an improved sensor circuit in a switching regulator circuit that includes a reduced temperature coefficient for the current limit in accordance with an embodiment of the present invention. 
     FIG. 3 is a schematic diagram illustrating an improved sensor circuit in a switching regulator circuit that includes a reduced temperature coefficient for the current limit in accordance with another embodiment of the present invention. 
     FIG. 4 is a schematic diagram illustrating an improved sensor circuit in a switching regulator circuit that includes a reduced temperature coefficient for the current limit in accordance with yet another embodiment of the present invention. 
     FIG. 5 is a schematic diagram illustrating an improved sensor circuit shown in further detail in accordance with another embodiment of the present invention. 
     FIG. 6 is a schematic diagram illustrating an improved sensor circuit shown in further detail in accordance with yet another embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     Throughout the specification, and in the claims, the term “connected” means a direct electrical connection between the things that are connected, without any intermediary devices. The term “coupled” means either a direct electrical connection between the things that are connected, or an indirect connection through one or more passive or active intermediary devices. The term “circuit” means one or more passive and/or active components that are arranged to cooperate with one another to provide a desired function. The term “signal” means at least one current signal, voltage signal or data signal. The meaning of “a”, “an”, and “the” include plural references. The meaning of “in” includes “in” and “on”. 
     The present invention relates to switching regulator and driver circuits. More particularly, the present invention relates to a method and apparatus that provides for a reduced temperature coefficient for the current limit sensor in a switching regulator/driver circuit. The present invention has determined that a need exists for an integrated circuit switching regulator/driver that eliminates the use of one or more “off-chip” resistors. By eliminating “off-chip” resistor elements, power consumption can be reduced and overall cost is reduced. The present invention utilizes an active element as the sense resistor (R S ) . The active element reduces the “chip area” that is necessary to provide the required sense resistor (R S ). The active sense resistor (R S ) eliminates the need for an external (“off-chip”) resistor. 
     FIG. 2 is a schematic diagram illustrating an example of a switching regulator sensor circuit ( 200 ) that is in accordance with the present invention. In FIG. 2, the switching regulator sensor circuit ( 200 ) includes a comparator circuit ( 210 ), a converter circuit ( 220 ), a sensor circuit ( 230 ), an output circuit ( 240 ), and a sense resistor (R SNS ). 
     Comparator circuit  210  includes a differential input having a sensor input (SNS) that is coupled to node N 20 , a reference input (REF) that is coupled to node N 21 , and an output that is coupled to node N 23 . A system output (Output) is coupled to node N 23 . Converter circuit  220  includes an output terminal (O) that is coupled to node N 21  and an input terminal (I) that is coupled to node N 22 . In one embodiment, converter circuit  220  is a trans-conductance cell. Sense resistor R SNS  is coupled between node N 21  and power supply node N PS20 . Sensor circuit  230  includes an input terminal (I) that is coupled to power supply node N PS20  and an output terminal (O) that is coupled to node N 20 . Sensor element  230  includes a resistance. Output circuit  240  includes a sense terminal (SNS) that is coupled to node N 20 , a power terminal (PWR) that is coupled to power supply node N PS20 , a driver terminal (DRV) that is coupled to node N 24 , and a ground terminal (GND) that is coupled to a circuit ground potential (GND). In one embodiment, output circuit  240  is a voltage-controlled current source. 
     In operation, a signal is coupled from a temperature independent reference voltage (V REF ) to node N 22 , and a supply voltage (V SUP ) is coupled to power supply node N PS20 . Converter circuit  220  produces a reference signal at node N 21  based on the signal received at node N 22 . The signal received at node N 21  is coupled to the reference input (REF) of comparator  210 . Sensor element  230  produces a signal at node N 20  in response to the power supply voltage (V SUP ) and a current (I S ). The signal received at node N 20  is coupled to the sensor input (SNS) of comparator  210 . The value of the current (I S ) is dependent on the output circuit ( 240 ). The value of the current (I S ) at node N 20  is dependent on the value of a signal received at the power terminal (PWR) in conjunction with a signal received at the drive terminal (DRV) of output circuit  240 . 
     Comparator  210  produces an output signal at node N 23  in response to the signals at nodes N 20 , N 21 . The voltage signal at the reference input (REF) is equal to the supply voltage (V SUP ) less the drop across the sense resistor (R SNS ). The voltage drop across sense resistor R SNS  is equal to the product of sensor current (I SNS ), flowing through the sense resistor (R SNS ), and the value of the sensor resistor (R SNS ). The value of the sensor current (I SEN ) is equal to the product of the temperature independent reference voltage (V REF ) and the value of the trans-conductance of converter circuit  220 . Therefore, the potential of the signal at node N 21 , is given by: 
     
       
           V   21   =V   SUP −( I   SNS   ·R   SNS ) 
       
     
     where: 
     
       
           I   SNS =( V   REF   ·g   M ), 
       
     
     and 
     
       
           V   21   =V   SUP −(( V   REF   ·g   M )· R   SNS ) 
       
     
     Similarly, the potential at node N 20  is determined by the supply voltage (V SUP ) and the voltage drop across the sensor element ( 230 ). Sensor element  230  has an associated resistance (R S ). The voltage drop across sensor circuit  230  is determined by the current (I S ) that is flowing through the sensor circuit ( 230 ), and the value of a resistance (R S  associated with the sensor circuit ( 230 ). Current I S  is determined by the output circuit ( 240 ). Therefore, the potential of the signal at node N 20 , is given by: 
     
       
           V   20   =V   SUP −( I   S   ·R   S ) 
       
     
     where I S  is proportional to the current flowing in the output circuit ( 240 ). 
     The output of the comparator circuit ( 210 ) changes logic states when the potential (V 20 ) of at node N 20  substantially reaches (or exceeds) the potential (V 21 ) at node N 21 . The peak value for the sensor current (I S ) occurs when potential V 20  equals potential V 21 . Thus, the sensor current (I S ) is determined by: 
     
       
         
           V 
           21 
           =V 
           20 
         
       
     
     
       
         (( V   REF   ·g   M )· R   SNS )=( I   S   ·R   S ) 
       
     
       I   S =(( V   REF   ·g   M )· R   SNS )/ R   S   
     where I S  is the peak sensor current. 
     Although V REF  is temperature independent, the various resistances (R SNS , R S ) and trans-conductance (g M ) by their very nature have temperature dependent resistances. The present invention seeks to control a current limit (I S ) that is temperature independent. 
     FIG. 3 is a schematic diagram illustrating an example of a switching regulator sensor circuit ( 300 ) that is in accordance with the present invention. In FIG. 3, the switching regulator sensor circuit ( 300 ) includes a comparator circuit ( 310 ), a trans-conductance cell ( 320 ), a voltage source ( 330 ), a driver circuit ( 335 ), an output circuit ( 240 ), a load (Z L ), and two resistors (R SNS1  and R SNS2 ). Output circuit  240  further includes two transistors (M 30 , M 31 ). 
     Comparator circuit  310  includes a non-inverting input (+) that is coupled to node N 31 , an inverting input (−) that is coupled to node N 30 , and an output terminal that is coupled to node N 34 . Trans-conductance cell  320  includes an output that is coupled to node N 31  and an input that is coupled to node N 32 . Voltage source  330  includes an output that is coupled to node N 32 . Resistor R SNS1  is coupled between node N 3 , and power supply node N PS30 . Driver circuit  335  includes an input that is coupled to node N 34  and an output that is coupled to node N 35 . 
     Resistor R SNS2  is coupled between node N 30  and power supply node N PS30 . Transistor M 31  includes a drain that is coupled to node N 30 , a gate that is coupled to node N 35 , and a source that is coupled to node N 33 . Transistor M 30  includes a gate that is coupled to node N 35 , a source that is coupled to node N 33 , and a drain that is coupled to power supply node N PS30 . In this configuration, transistor M 30  is referred to as a “switch transistor.” Load Z L  is coupled between node N 33  and power supply node N PS31 . A circuit ground potential (GND) is coupled to power supply node N PS30 . 
     The components of FIG. 3 function similarly to like named components in FIG.  2 . Driver circuit  335  is arranged to drive transistors M 30  and M 31  when the circuit is configured to operate as a regulator. Transistor M 30  represents an output driver that drives current into load Z L , while transistor M 31  (sharing a common gate and source with transistor M 30 ) provides a current (I S ) that is representative of the current that is driven into load Z L . Transistor M 31  and resistor R SN52  operate as a current sense circuit that provides a voltage corresponding to the current that is driven into load Z L . 
     Comparator circuit  310  changes the logic state of its output when an over current condition occurs. The over current condition may be used to shut down the driver circuit ( 335 ) such that transistors M 30  and M 31  are disabled. By disabling transistors M 30  and M 31 , the output current that is delivered to load Z L  is limited. 
     The over current condition is detected by comparator circuit  310  when the potential at node N 30  equals (or exceeds) the potential at node N 31 . The potential of at node N 31  is obtained utilizing the same method employed in deriving the value of the voltage present at node N 21  of FIG.  2 . Therefore, the potential of the signal (V + ) at the noninverting input (+) of comparator circuit  310  is: 
     
       
         
           V 
           + 
           =V 
           SUP 
           −V 
           SNS1 
         
       
     
     where V SNS1  represents the voltage drop across resistor R SNS1 . 
     
       
           V   +=   V   SUP −( I   SNS   ·R   SNS1 ) 
       
     
     where I SNS =(V REF ·g M ) 
     
       
           V   +   =V   SUP −[( V   REF   ·g   M )· R   SNS1 ] 
       
     
     where V REF  is the voltage signal produced by voltage source  330 . 
     In one example, voltage source  330  is a band gap type of reference circuit. Voltage source  330  may also be derived from another circuit. Trans-conductance cell  320  and voltage source  330  may be combined into a single circuit. Also, other circuitry (not shown) in the switching regulator or driver circuit may be utilized to provide the voltage at node N 32 . 
     The potential of node N 30  is obtained utilizing a similar method employed in deriving the potential at node N 20  of FIG.  2 . The potential of node N 30  (V − ) is equal to the difference of a supply voltage (V SUP ) and the voltage drop (V SNS2 ) across resistor R SNS2 . The voltage drop across resistor R SNS2  is equal to the product of the current (I S ) flowing in resistor R SNS2  and resistance of the resistor (R SNS2 ). In one embodiment, the resistance of resistor R SNS2  is much greater than the resistance across the drain-source (R DSON(M31) ) of M 31  when the transistor is “on” (i.e., at least ten times greater). The current (I S ) flowing in transistor M 31  is proportional to an output circuit current (I SW ) flowing in transistor M 30 . A further property of this embodiment is that the potential across the drain-source (R DSON(M30) ) of transistor M 30  is approximately equal to the potential across resistor R SNS2 . 
     The value of the load (Z L ) and a driver voltage (V DRV ) applied to the gates of the transistors (M 30  and M 31 ) determines the value of the output circuit current (I SW ). Changes in either the load (Z L ) or the driver voltage (V DRV ) cause a corresponding change in the potential (V − ) at node N 30 . When the voltage signal levels of the differential inputs of comparator  310  are approximately equal, the comparator circuit ( 310 ) will trip. Hence, the output circuit can be determined by: 
     
       
         
           V 
           + 
           =V 
           − 
         
       
     
     Substituting from the above equations and text results in 
     
       
           V   SUP −( I   SNS   ·R   SNS )= V   SUP −[( R   DSON(M30) )· I   SW ] 
       
     
     
       
         ( I   SNS   ·R   SNS )=( R   DSON(M30) ) ·I   SW   
       
     
     where 
     
       
         
           I 
           SNS 
           =V 
           REF 
           ·g 
           M 
         
       
     
     Further substitutions result in 
     
       
         ( V   REF   ·g   M ) ·R   SNS =( R   DSON(M30) ) ·I   SW   
       
     
     Solving for the output circuit current (I SW ) yields 
     
       
           I   SW   =V   REF   ·g   M   ·R   SNS /( R   DSON(M30) ) 
       
     
     For simplicity, the trans-conductance value of g M  can be represented as an equivalent resistance value R EQ , where R EQ =1/g M . Thus, I SW  is given by: 
     
       
           I   SW   =V   REF ·( R   SNS )/( R   DSON(M30) ) ·R   EQ   
       
     
     When making an allowance for temperature considerations, each resistance includes its inherent resistance value (R) plus a temperature coefficient (α·T). This resistance is represented as: 
     
       
           R ( T )= R   0 ·(1+α R   ·T ) 
       
     
     giving: 
     
       
           R   SNS ( T )= R   SNS ·(1+α SNS   ·T ) 
       
     
     
       
           R   DSON ( T )= R   DSON ·(1+α DSON   ·T ) 
       
     
     
       
           R   EQ ( T )= R   EQ ·(1+α EQ   ·T ) 
       
     
     Substituting for a temperature dependent equation yields 
     
       
           I   SW ( T )= V   REF   ·[R   SNS ·(1+α SNS   ·T )/ R· (1+α EQ   ·T )·( R   DSON )·(1+α DSON   ·T )] 
       
     
     Linearizing the equation yields 
     
       
           I   SW ( T )= V   REF ·[( R   SNS )/ R   EQ ·( R   DSON )]·[1+(α SNS −α EQ −α DSON )· T]   
       
     
     To obtain a temperature independent output current (I SW ), the values of the sum of the temperature coefficients (α) should equal zero or equal a value as close to zero as possible. 
     
       
         (α SNS −α EQ −α DSON )=0 
       
     
     The temperature coefficient values (α) are determined by the materials used to construct the resistors (e.g., passive device), or the type of materials in the active device (e.g., equivalent resistance). For example, a suitable coefficient for R DSON  when utilizing a MOSFET type device could be 5000 ppm/° C. A suitable coefficient for R SNS  when utilizing a well type resistor could be 6000/° C. Similarly, a suitable coefficient for R EQ  (the equivalent resistance of the trans-conductance cell circuit ( 320 )) when utilizing a heavily doped resistor could be 1000/° C. In one embodiment, complementary material types are used for the resistors in the above description such that the value of the difference of temperature coefficients of the two sensor resistors (α SEN −α R ) is close to or equal to the temperature coefficient value of the “switch” transistor (α DSON ). In such an embodiment, the output circuit current (I SW ) temperature coefficient could be close to zero. 
     FIG. 4 is a schematic diagram illustrating an example of another switching regulator sensor circuit ( 400 ) that is in accordance with the present invention. In FIG. 4, the switching regulator sensor circuit ( 400 ) includes a comparator circuit ( 410 ), a trans-conductance cell ( 420 ), a temperature independent voltage source ( 430 ), an output circuit ( 240 ), a load (Z L ), and two resistors (R SNS1 , R SNS2 ). Output circuit  240  further includes two transistors (M 40 , M 41 ) 
     Comparator circuit  410  includes a non-inverting input (+) that is coupled to node N 40 , an inverting input (−) that is coupled to node N 41 , and an output that is coupled to node N 44 . Trans-conductance cell  420  includes an output that is coupled to node N 41  and an input that is coupled to node N 42 . Temperature independent voltage source  430  includes an output that is coupled to node N 42 . Resistor R SNS1  is coupled between node N 41  and power supply node N PS41 . Resistor R SNS2  is coupled between node N 40  and power supply node N PS41 . Transistor M 41  includes a drain that is coupled to node N 40 , a gate that is coupled to node N 45 , and a source that is coupled to node N 43 . Transistor M 40  includes a gate that is coupled to node N 45 , a source that is coupled to node N 43 , and a drain that is coupled to power supply node N PS41 . Load Z L  is coupled between node N 43  and power supply node N PS40 . A supply voltage (V SUP ) is coupled to power supply node N PS40 . A circuit ground potential (GND) is coupled to power supply node N PS41 . 
     The configuration of the switching regulator sensor circuit ( 400 ) of FIG. 4 functions similarly to the switching regulator sensor circuit ( 300 ) of FIG.  3 . The components of FIG. 4 function similarly to like named components in FIG.  3 . In operation, the transistors (M 40 , M 41 ) of switching regulator sensor circuit ( 400 ) are p-type transistors in contrast to the n-type transistors shown in the other switching regulator sensor circuit ( 300 ). Due to the inherent properties of the active devices used in FIGS. 3 and 4, the temperature coefficient equation for switching regulator sensor circuit  400  is substantially the same as the equation derived above for switching regulator sensor circuit  300 . The temperature coefficient equation utilized to obtain a temperature independent current for switching regulator sensor circuit  400  is: 
     
       
         (α SNS −α EQ −α DSON )=0 
       
     
     The temperature coefficient values (α) are determined by the materials used to construct the resistors, or the type of materials used to construct the active devices (e.g., transistors M 40 , M 41 ). 
     FIG. 5 illustrates another embodiment of the present invention. Like components from FIGS. 3 and 5 are labeled identically. FIG. 5 is a schematic diagram illustrating one embodiment of a switching regulator sensor circuit ( 500 ). Switching regulator sensor circuit  500  includes a comparator circuit ( 310 ), a trans-conductance cell ( 320 ), a temperature independent voltage source ( 330 ), an output circuit ( 240 ), a load (Z L ), and two resistors (R SNS1  and R SNS2 ). Trans-conductance cell circuit  320  further includes an operational amplifier ( 510 ), a transistor (M 50 ), and a resistor (R 50 ). Output circuit  240  further includes two transistors (M 30 , M 31 ). Refer to FIG.  3  and the related discussion for the overall operation and connections of the like designated components. 
     Operational amplifier  510  includes a non-inverting input (+) that is coupled to node N 50 , an inverting input (−) that is coupled to node N 51 , and an output that is coupled to node N 52 . Transistor M 50  includes a gate that is coupled to node N 52 , a source that is coupled to node N 51 , and a drain that is coupled to node N 53 . Resistor R 50  is coupled between node N 51  and power supply node N PS51 . Load Z L  is coupled between node N 53  and power supply node N PS50 . A supply voltage (V SUP ) is coupled to power supply node N PS50 . A circuit ground potential (GND) is coupled to power supply node N PS51 . 
     In operation, a reference signal is coupled from temperature independent reference voltage ( 330 ) to node N 32 . In one embodiment, temperature independent reference voltage ( 330 ) is a band gap reference. Operational amplifier  510  compares the reference signal to a feedback signal at node N 51  and produces an output at node N 52  based on the comparison. Transistor M 50  produces a current (I SNS ) in response to the potential at node N 52  and the potential at node N 51 . The current (I SNS ) flows through the resistor (R 50 ) to produce the feedback signal. When the potential at node N 32  and node N 51  are approximately equal, the current (I SNS ) is approximately given by: 
     
       
         
           I 
           SNS 
           =V 
           REF 
           /R 
           50 
         
       
     
     In this way, trans-conductance cell  320  functions as a voltage to current converter. Resistor R 50  operates similar to R EQ  previously described with respect to FIG.  3 . 
     FIG. 6 illustrates another embodiment of the present invention. Like components from FIGS. 4 and 6 are labeled identically. FIG. 6 is a schematic diagram illustrating an embodiment of a switching regulator sensor circuit ( 600 ). Switching regulator sensor circuit  600  includes a comparator circuit ( 410 ), a trans-conductance cell circuit ( 420 ), a temperature independent voltage source ( 430 ), an output circuit ( 240 ), a load (Z L ), and two resistors (R SNS1 , R SNS2 ). Trans-conductance cell circuit  420  further includes an operational amplifier ( 610 ), a transistor (M 60 ), and a resistor (R 60 ). Output circuit  240  further includes two transistors (M 40 , M 41 ). Refer to FIG.  4  and the related discussion for the overall operation and connections of the like designated components. 
     Operational amplifier  610  includes a non-inverting input (+) that is coupled to node N 60 , an inverting input (−) that is coupled to node N 61 , and an output that is coupled to node N 62 . Transistor M 60  includes a gate that is coupled to node N 62 , a drain that is coupled to node N 61 , and a source that is coupled to node N 63 . Resistor R 60  is coupled between node N 61  and power supply node N PS60 . Load Z L  is coupled between node N 63  and power supply node N PS61 . A supply voltage (V SUP ) is coupled to power supply node N PS60 . A circuit ground potential (GND) is couple to power supply node N PS61 . 
     The configuration of the switching regulator sensor circuit ( 600 ) of FIG. 6 functions similarly to the switching regulator sensor circuit ( 500 ) of FIG.  5 . The components of FIG. 6 function similarly to like named components in FIG.  5 . In operation, the active components of switching regulator sensor circuit ( 600 ) are p-channel type in contrast to the n-channel type of the switching regulator sensor circuit ( 500 ). 
     Although active circuits, such as, voltage-controlled current sources, trans-conductance cells, comparators, etc include FET&#39;s in the above description, it is understood and appreciated that other active devices could be used as well. For example, NPN transistors, PMOS transistors, MOSFET&#39;s, NMOS transistors, GaAs FET&#39;s, JFET&#39;s, Darlington pairs, bipolar junction transistors, as well as others may be used in the switching regulator sensor circuit. An important design criteria is that the temperature coefficient (α) be amenable to cancellation when utilized in the above equation or another equation derived from the use of the aforementioned device type. 
     The above specification, examples and data provide a complete description of the manufacture and use of the composition of the invention. Since many embodiments of the invention can be made without departing from the spirit and scope of the invention, the invention resides in the claims hereinafter appended.