Abstract:
The present invention provides a method and apparatus  200  for phase tracking coherent detection in a communication system. The method provides receiving a waveform which carries information including both phase reference information and data information. The received waveform is phase-rotated by a phase offset estimate. Phase reference information and data information is extracted from the phase-rotated received waveform. A phase offset estimate is calculated based on the extracted phase reference information, extracted data information, and the received waveform. The apparatus  200  provides a phase rotator  206  for phase-rotating an received waveform by an input phase offset estimate. A data detector  210  is provided which extracts phase reference information and data information from the phase-rotated version of the received waveform. A phase correction angle calculator  214  is provided to calculate the phase offset estimate based on the extracted phase reference information, the extracted data information, and the received waveform.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     Not Applicable. 
     STATEMENT REGARDING FEDERALLY SPONSORED RESEARCH OR DEVELOPMENT 
     Not applicable. 
     BACKGROUND OF THE INVENTION 
     The present invention generally relates to coherent detection. More specifically, the present invention relates to tracking the absolute phase of a received waveform for coherent detection in a phase or frequency modulated communication system where the transmitted waveform consists of both phase reference symbols and data symbols. 
     Phase modulation (“PM”) is widely used in communication systems. In phase modulation schemes, data is represented by either the absolute phase of the waveform or by changes in the phase of the waveform. One reason for the popularity of PM is the robustness of PM with respect to additive white Gaussian noise. Common examples include Binary Phase Shift Keying (“BPSK”), Quadrature Phase Shift Keying (“QPSK”), and Gaussian Minimum Shift Keying (“GMSK”). QPSK, for example, represents two bits of information (which may assume a total of four different values) in every symbol. A symbol consists of a phase shift by one of four possible phase shift values. The phase shift values are typically chosen to be plus/minus 45 degrees and plus/minus 135 degrees. 
     As the demand for communication bandwidth rises, the concern over efficient use of available bandwidth similarly rises. GMSK has been chosen by many because of its relatively efficient use of bandwidth. In GMSK, symbols are represented by gradual changes in phase which result in a power spectral density that rapidly falls off. This allows GMSK channels to be packed relatively efficiently into a given frequency band. 
     The scenario of interest involves modulation schemes that map information to the absolute phase of the waveform. As such, phase shift values are measured with respect to some reference. In order for a receiver to extract data from the received waveform, the phase shift values relative to the reference must be known. A receiver, therefore, needs to have knowledge of the phase reference. The transmitted waveform may contain symbols whose express purpose is to provide the receiver with explicit phase reference. For example, the GMSK waveform, as defined in the Advanced EHF Waveform Functional Description includes a specification for transmitting both phase reference symbols and data symbols. A receiver may then extract the absolute phase directly from the phase reference symbols. 
     Typically, phase lock loops or similar circuits with local oscillators are utilized in the process of signal channelization and detection. In some cases, the local oscillators may be used to provide phase references to data detection circuitry. Data detection circuitry may even be able to assert control over the local oscillators once more is known about the true phase reference. However, access to local oscillators may not always be practical. In some communications systems, the local oscillators may be set to specific frequencies or may be too slow to react to incoming phase reference information. In addition, the local oscillators may be used exclusively to extract the baseband waveform from an intermediate frequency, while the task of extracting both phase reference and data decisions from the baseband waveform falls on independent circuitry. To this end, methods have been developed to determine the absolute phase of the received waveform based on phase reference symbols. However, the methods that have been developed thus far focus on determining the absolute phase using only the phase reference symbols. Such methods fail to take advantage of additional phase reference information that may be obtained from the data symbols, and thus do not track the absolute phase of the received waveform as well as they potentially could. 
     A need exists for a coherent detection system that is capable of achieving a better phase reference than systems which utilize only phase reference symbols. 
     SUMMARY OF THE INVENTION 
     It is an object of the present invention to provide an improved scheme for tracking the absolute phase of a received waveform in a communication system. 
     It is another object of the present invention to provide an improved scheme for tracking the absolute phase of a received waveform in a coherent detection system which utilizes both phase reference symbols and data symbols contained in the waveform. 
     It is still another object of the present invention to provide an improved scheme for tracking the absolute phase of a received waveform in a coherent detection system in which a nominal waveform is generated from the detected data and is fed back and compared to the received waveform to determine the necessary amount of phase correction. 
     A preferred embodiment of the present invention provides a method and apparatus for coherent detection including tracking the phase of a received waveform containing both phase reference symbols and data symbols. The received waveform is phase-rotated by a phase offset estimate. Data decisions are extracted from the phase-corrected version of the received waveform. A nominal waveform is generated based on the data decisions resulting from the phase-corrected version of the received waveform. A phase measurement is generated from the difference in phase between the received waveform and the nominal waveform. The received waveform is phase-rotated by the phase offset estimate which is calculated from the phase measurements. 
     The apparatus provides a phase rotator for phase-rotating the received waveform by the phase offset estimate. A data detector is provided which extracts data decisions from the phase-corrected version of the received waveform. A waveform modulator is provided to generate a nominal waveform based on the data decisions generated from the data detector. A buffer is provided to time-align the received waveform with the nominal waveform. A phase error signal calculator is provided to generate a measurement of the difference between the phase of the nominal waveform and the phase of the received waveform. A phase correction angle calculator is provided to estimate the phase offset from the phase measurements. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 illustrates a phase tracking coherent detection system utilizing linear interpolation of the phase reference measurements provided by the phase reference symbols to establish a phase reference estimate. 
     FIG. 2 illustrates a high level phase tracking coherent detection system according to a preferred embodiment of the present invention. 
     FIG. 3 illustrates a phase tracking coherent detection system according to a preferred embodiment of the present invention. 
     FIG. 4 illustrates a GMSK phase tracking coherent detection system according to an embodiment of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     One method for extracting the absolute phase from a received waveform is to use linear interpolation of phase measurements generated from phase reference symbols interspersed throughout the transmission. FIG. 1 illustrates an example of a GMSK data detector  100  with a linear interpolated phase reference estimator. Reference symbols may be sprinkled throughout the transmitted sequence in a predetermined manner. For this example, there is a reference symbol sequence at the beginning and end of each block of data symbols. The reference ROM  102  stores the bit sequences corresponding to the known patterns of the phase reference symbols. Before the arrival of the data block, the GMSK modulator  104  uses the reference ROM  102  output to generate the expected pattern of phase reference symbols. The conjugate of the expected pattern of phase reference symbols is generated by the complex conjugator  106 . Note that FIG. 1 is used to describe the correct functional flow, and is not intended to suggest a hardware realization. The complex conjugator  106  output is multiplied by the received waveform by the multiplier block  108 . The results of the complex multiplication constitute a set of phase offset measurements. The phase offset measurements are input to the linear interpolator  110 . The linear interpolator  110  computes a phase offset estimate from the set of phase offset measurements. At the end of the data block, a second received pattern of phase reference symbols is compared to a second expected pattern of phase reference symbols. As before, the comparison results in a phase offset estimate. Given a phase offset estimate at both the beginning and end of the data block, the linear interpolator  110  calculates a phase offset estimate for the duration of the data block. While the linear interpolation of the phase reference estimate is being calculated for a data block, the received waveform is delayed by the buffer  112  to allow the linear interpolator time to make its estimated phase correction calculation. The delayed version of the received waveform is derotated by multiplying it by the complex conjugate of the linearly estimated phase during the duration of the data block. The multiplication is performed by the multiplier  114 . The delayed and derotated waveform output from the multiplier  114  is then sent to a GMSK data detector  116  which generates the data decisions. 
     The linear interpolation method just discussed explicitly calculates a phase correction at the beginning and at the end of each data block where phase reference symbols reside. However, it only provides a linearly interpolated estimate of the phase correction. In some cases this linearly estimated phase correction is adequate. However, we can achieve superior performance at a small cost to implementation complexity. 
     FIG. 2 illustrates a high level phase tracking coherent detection system  200  according to a preferred embodiment of the present invention. The received waveform  202  and the phase offset estimate  204  are input to a phase rotator  206 . The purpose of the phase rotator  206  is to phase rotate the input received waveform  202  by the phase offset estimate  204 , thereby providing a phase-corrected version of the received waveform  202 . The phase-corrected waveform  208 , output from the phase rotator  206 , is input to a data detector  210 . 
     The purpose of the data detector  210  is to generate data decisions from the phase-corrected waveform  208 . The output  212  of the data detector  210  contains data decisions, which may take the form of a stream of bits. In the specific case of a GMSK data detector, the output  212  of the data detector  210  may take the form of a stream of bits (or chips) comprising phase reference bits and data bits. The data decisions from the output  212  of the data detector  210  are forwarded to some external receiving entity. The data decisions from the output  212  of the data detector  210  are also fed back to an input of the phase correction angle calculator  214 . 
     The phase correction angle calculator  214  completes the data decision feedback loop by calculating the phase offset estimate  204  discussed above. The phase correction angle calculator  214  also receives the received waveform  202  as input. The phase offset estimate  204  is calculated based on the received waveform  202  and the data decisions fed back from the output  212  of the data detector  210 . 
     A more detailed illustration of a phase tracking coherent detection system  300  according to a preferred embodiment of the present invention is illustrated in FIG.  3 . The phase correction angle calculator  214  of FIG. 2 corresponds to the phase correction angle calculator  328  of FIG.  3 . The received waveform  302  and the offset estimate  304  are input to a phase rotator  306 . The purpose of the phase rotator  306  is to phase rotate the received waveform  302  by the phase offset estimate  304 , thereby providing a phase-corrected version of the received waveform  302 . The phase rotator  306  preferably takes the form of a simple multiplier. The phase-corrected waveform  308 , output from the phase rotator  306 , is input to a data detector  310 . 
     The purpose of the data detector  310  is to generate the data decisions from the phase-corrected waveform  308  applied to the input of the data detector  310 . The output  312  of the data detector  310  contains data decisions which may take the form of a stream of bits (or chips). In the specific case of a GMSK data detector, the output of the data detector  310  preferably takes the form of a stream of bits comprising phase reference bits and data bits. The data decisions from the output  312  of the data detector  310  are forwarded to the next receiving entity. The data decisions from the output  312  of the data detector  310  are also fed back to an input of a waveform modulator  314 . 
     The purpose of the waveform modulator  314  is to generate the nominal waveform that should have been received based on the data decisions generated by the data detector  310 . If the received data decisions were made perfectly, the nominal waveform  316  from the waveform modulator  314  would be identical to the transmitted waveform. For the preferred embodiment illustrated in FIG. 3, the nominal waveform  316  is generated by the waveform modulator  314  in the angle domain. For the specific case of a GMSK modulator, the data decisions include both phase reference chips and data chips, and the output  316  of the waveform modulator  314  would be the nominal GMSK waveform for the data detector input. The output nominal waveform  316  from the waveform modulator  314  is applied to an input of an error signal generator  318 . 
     The received waveform  302  is converted to the angle domain by an I-Q to angle converter  319  and applied to the input of a buffer  320 . The received waveform  302  is ultimately compared with the nominal waveform  316 . However, there is a time delay between the time of arrival of the received waveform  302  and the time of generation of the nominal waveform  316 . In order for an appropriate comparison to be made between the received waveform  302  and the nominal waveform  316 , the two waveforms must be properly aligned in time. To this end, the buffer  320  is utilized to output a time-aligned version of the received waveform in the angle domain  322  to an input of the error signal generator  318 . 
     The error signal generator  318  accepts as input the nominal waveform  316  from the waveform modulator  314  and the time-aligned angle domain version of the received waveform  322  from the buffer  320 . In the preferred embodiment illustrated in FIG. 3, the error signal generator  318  computation is performed in the angle domain. Thus, the error signal generator  318  may take the form of a simple modulo subtracting element. In the case of a simple subtracting element, a perfect match between the nominal waveform  316  and the time-aligned angle domain version of the original waveform  322  would result in a zero output error signal  324 . The phase offset measurement  324  output from the error signal generator  318  is applied to the input of a phase correction element  326 . 
     For illustrative purposes, an alternate embodiment of a coherent detection system  400  according to the present invention is shown in FIG.  4 . The received waveform  402  is received in the I-Q domain. The received waveform  402  is time-aligned by a delay element  420  and complex conjugated by a complex conjugator  417 . The waveform modulator  414  outputs a nominal waveform  416  in the I-Q domain. The error signal generator  418  accepts as input the nominal waveform  416  from the waveform modulator  414  and the complex conjugated time-aligned version of the received waveform  402 . Since the error signal generator computation is being performed in the I-Q domain, versus the angle domain computation in the preferred embodiment illustrated in FIG. 3, the error signal generator  418  may take the form of a multiplication element. 
     Referring back to the preferred embodiment illustrated in FIG. 3, the purpose of the phase correction element  326  is to make a phase offset estimate decision based on the phase offset measurement  324  from the error signal generator  318 . The phase offset estimate  304  is then output to the phase rotator  306 . The phase correction element  326  may take many forms. In one specific case, the phase correction element  326  may calculate the phase offset estimate  304  based on a weighted averaging scheme. The weighted averaging scheme may be an information-type-based weighted averaging scheme where the input phase offset estimate  324  is weighted with a first weight during periods of the waveform containing known reference symbols, and where the input phase offset measurement is weighted with a second weight during periods of the waveform containing data symbols. Optionally, the weighted averaging scheme may be a waveform-quality-based weighted averaging scheme where the weights used in calculating the weighted average are real-time variables and are adjusted in real-time based on signal quality measurements (e.g. signal-to-noise ratio). When the signal quality is poor, perhaps due to natural or manmade interference, the weights may be reduced. The phase correction element  326  may take the form of a common filter, such as a Kalman filter. Completing the data decision feedback loop, the phase offset estimate  304  output from the phase correction element  326  is applied to an input of the phase rotator  306 . 
     The phase correction action performed by the data decision feedback loop continues for the duration of the received waveform  302 . 
     In the discussion of the phase tracking demodulation system  300  according to a preferred embodiment of the present invention illustrated in FIG. 3, information passed between elements of the figure was named in general descriptive terms. The exact form of the information may vary. In a preferred embodiment of the present invention the original carrier  302 , phase offset estimate  304 , phase-corrected version of the received waveform  308 , nominal waveform  316 , time-aligned angle domain version of the received waveform  322 , phase offset measurement  324 , and the data decisions output from the data detector  310  take the form of a digital signals. 
     While particular elements, embodiments and applications of the present invention have been shown and described, it will be understood that the invention is not limited thereto since modifications may be made by those skilled in the art, particularly in light of the foregoing teachings. It is therefore contemplated by the appended claims to cover such modifications as incorporate those features which come within the spirit and scope of the invention.