Abstract:
A method for measuring the waveform of light is provided, which makes it possible to synchronize easily the phase of sampling light with the phase of target light even if the target light is ultra-high speed pulsed light and is transmitted by way of long transmission channel, and to measure the waveform of target light with sufficient time resolution in real time. The method comprises the steps of: (a) generating sampling light having a pulse width sufficiently narrower than that of the target light from the target light; a repetition frequency of the sampling light having a constant difference with respect to a repetition frequency of the target light; (b) supplying the sampling light and the target light to a nonlinear optical member to generate cross-correlated light between the sampling light and the target light; and (c) measuring waveform of the target light based on the cross-correlated light. Preferably, the step (a) comprises the substeps of (a- 1 ) extracting clock light from the target light; the clock light being synchronized with the target light; (a- 2 ) generating an electrical clock signal from the clock light; (a- 3 ) generating an electrical driving signal in such a way that a constant frequency difference exists between a frequency of she electrical driving signal and that of the electrical clock signal; and (a- 4 ) generating the sampling light based on the electrical driving signal.

Description:
BACKGROUND OF THE INVENTION  
         [0001]    1. Field of the Invention  
           [0002]    The present invention relates to optical measurement of the waveform of target light and more particularly, to a method of measuring the waveform of target light and an apparatus for measuring the same, which are applicable to measurement of the waveform of ultra-high speed pulsed light used for optical communication and/or optical information processing.  
           [0003]    2. Description of the Related Art  
           [0004]    In recent years, the capacity of data to be transmitted in optical communications systems has been increasing rapidly and accordingly, not only the techniques for the wavelength multiplexing method that transmits the data using different wavelengths of signal light but the techniques for raising the data transmission rate in each wavelength to 100 Gb/s or higher have been being researched and developed actively. Under such circumstances, there have been the increasing need to develop the techniques for generating stable, coherent, ultra-high speed optical pulses and to measure the waveform of the ultra-high speed optical pulse train in real time with sufficiently high time resolution. In particular, the “eye pattern measurement” that measures directly an optical pulse train modulated by random bit data is essential to evaluate the characteristics of optical transmission systems.  
           [0005]    A typical one of the known methods of measuring optical pulse trains is to use a ultra-high speed photoelectric converter and an electrically sampling oscilloscope. In this case, the “eye pattern measurement” can be performed, but in the present circumstances, the higher end of the measurable frequency range of light is, at most, approximately 40 GHz. As a result, it is difficult to measure the waveform of ultra-high speed optical pulse trains having a data transmission rate that exceeds about 40 Gb/s in each wavelength in real time with sufficiently high time resolution.  
           [0006]    To solve the above-described difficulty, a method of measuring the waveform of target light has been developed and actually used. In this method, pulsed target light to be measured and pulsed sampling light having a sufficiently narrower pulse width than the target light is supplied to a specific nonlinear optical member, thereby generating intensity cross-correlated light between the target light and the sampling light due to nonlinear optical effects. On the basis of the cross-correlated light thus generated, the waveform of the target light is measured. In this method, the target light can be optically sampled and therefore, the-above-described difficulty can be solved. Specifically, the waveform of ultra-high speed optical pulses having a data transmission rate that exceeds about 40 Gb/s in each wavelength can be measured in real time with sufficiently high time resolution.  
           [0007]    Examples of the prior-art apparatuses of this type for measuring the waveform of target light pulses using the above-described method are disclosed in the Japanese Non-Examined Patent Publication No. 8-29814 published in 1996 and the Japanese Non-Examined Patent Publication No. 9-160082 published in 1997. FIG. 1 shows a typical one of the prior-art apparatuses of this type, in which thick lines with arrows indicate the flow of optical signals while thin lines with arrows indicate the flow of electrical signals.  
           [0008]    The prior-art measuring apparatus  200  comprises a driving signal oscillator  262 , a sampling light source  263 , a nonlinear optical member  264 , an optical filter  265 , an optical detector  266 , an electrical signal processing circuit  267 , and a display device  268 . The apparatus  200  itself is electrically and optically connected to an external apparatus  261 .  
           [0009]    The external apparatus  261  includes a driving signal oscillator  271  that oscillates an electrical driving signal SD 1  with a frequency f O  and a target light source  272  that is driven by the oscillator  271  to emit pulsed target light LT O . The target light LT O  thus emitted has a repetition frequency equal to the frequency f O  of the driving signal SD 1 . An example of the waveform of the target light LT O  is shown by the waveform a in FIG. 14.  
           [0010]    The oscillator  262 , which is electrically connected to the oscillator  271  provided in the external apparatus  261 , oscillates a driving signal SD 2  having a frequency f S  synchronized in phase with the driving signal SD 1  having the frequency f 0 . The reason why the oscillator  262  is electrically connected to the oscillator  271  is to synchronize the phase of the target light LT O  with the phase of the sampling light LT S . Because of the phase synchronization between the light LT O  and LT S , the fluctuation of time difference δt of each pulse of the target light LT O  from each pulse of the sampling light LT S , (i.e., mutual jitter), is decreased. Thus, the time resolution can be prevented from degrading. In principle, possible time resolution is approximately equal to the pulse width of the sampling light LT S .  
           [0011]    The sampling light source  263  is driven by the driving signal oscillator  262 , emitting the pulsed sampling light LT S . The sampling light LT S  thus emitted has a repetition frequency f S , where f S =(f O /N)−Δf, f O  is the repetition frequency of the target light LT O , Δf is a frequency difference, and N is a natural number (i.e., N=1, 2, 3, 4, . . .). The repetition frequency f S  of the sampling light LT S  is slightly different by Δf from the divided frequency of the target light LT O  by N, i.e., (f O /N). For example, when N=1, the sampling light LT S  has a waveform b shown in FIG. 14. In this case, each pulse of the sampling light LT S  has a time difference δt from the corresponding pulse of the target light LT O .  
           [0012]    The target light LT O  and the sampling light LT S  thus generated enters the nonlinear optical member  264 , emitting intensity cross-correlated light LT CC  between the light LT O  and LT S  thus supplied.  
           [0013]    The nonlinear optical member  264  may be made of a ferroelectric crystal such as KTP (KTiOPO 4 ) that causes a secondary nonlinear optical effect, e.g., the Sum Frequency Generation (SFG). Alternately, the member  264  maybe formed by a semiconductor optical amplifier or a quartz-system optical waveguide such as an optical fiber that causes a tertiary nonlinear optical effect, e.g., the Four Wave Mixing (FWM) . The member  264  is used to emit the intensity cross-correlated light LT CC  between the pulses of the target light LT O  and the sampling light LT S . For example, the cross-correlated light LT CC  thus emitted has a waveform shown by the waveform c in FIG. 14. The cross-correlated light LT CC  has a repetition frequency equal to the repetition frequency f S  of the sampling light LT S .  
           [0014]    Here, the time difference δt of the pulse of the sampling light LT S  from the corresponding pulse of the target light LT O  corresponds to the sampling time. Thus, it is expressed by the following equation (1).  
               δ                 t     =         1     f   s       -     N     f   0         ≅       Δ                 f       f   s   2                 (   1   )                               
 
           [0015]    For example, when the repetition frequency f S  of the sampling light LT S  is set as 1 GHz and the frequency difference Δf is set as 100 kz, the time difference δt is given as 0.1 ps (picosecond) by the equation (1).  
           [0016]    The optical filter  265  removes the target light LT O  and the sampling light LT S  and their secondary and higher harmonics (which serve as background light LT B  of the intensity cross-correlated light LT CC ), allowing only the cross-correlated light LT CC  to pass through the filter  265 .  
           [0017]    The optical detector  266  photoelectrically converts the cross-correlated light LT CC  thus passed through the filter  265  to generate a pulsed electrical signal S CC . The signal S CC  is supplied to the signal processing circuit  267 .  
           [0018]    The detector  266  needs to have a frequency band equal to or higher than the repetition frequency f S  of the sampling light LT S . This is due to the fact that each pulse of the cross-correlated light LT CC  needs to be photoelectrically converted separately in such a way as to cause no interference with its adjoining pulses in order to display correctly an eye pattern shown by the waveform d in FIG. 14 in the display device  268 . However, for example, even if the repetition frequency f O  of the target light LT O  is 100 GHz, the repetition frequency f S  of the sampling light LT S  can be lowered to approximately 100 MHz by setting the dividing factor N as 1000 in the sampling light source  263 . Thus, it is sufficient for the optical detector  266  to have a frequency band of approximately 100 MHz.  
           [0019]    The electrical signal processing circuit  267  samples the peak value of the pulsed electrical signal S CC  outputted by the detector  266  in synchronization with the repetition (i.e., sampling) frequency f S  of the sampling light LT S . Thus, the circuit  267  generates an electrical peak signal S CCP  that represents the peak values of the electrical signal S CC  and supplies the same to the display device  268 . In FIG. 14, the peak values of the electrical signal S CC  are shown by circular dots or spots in the vicinity of the waveform c.  
           [0020]    The display device  268  displays the waveforms so as to be overlapped with each other on its screen at a period of [1/(N·Δf)] on the basis of the electrical peak signal S CCP  outputted by the signal processing circuit  267 . Thus the device  268  displays the eye pattern as shown by the waveform d.  
           [0021]    The measuring person or tester evaluates the characteristics of the optical transmission system according to the opening level of the eye pattern displayed on the screen of the display device  263 .  
           [0022]    As explained above, with the prior-art measured apparatus  200  shown in FIG. 1, to synchronize the phase of the sampling light LT S  with the phase of the target light LT O , the driving signal oscillator  271  provided in the external apparatus  261  is electrically connected to the driving signal oscillator  262  provided near the sampling light source  263 . However, for example, when the ultra-high speed pulses of the target light LT O  having a repetition frequency that exceeds 40 Gb/s, which has been emitted in the external apparatus  261  and transmitted therefrom by way of an optical transmission path (e.g., optical fibers) of several kilometers in length, is measured in real time, it is not realistic to electrically interconnect the oscillator  262  with the oscillator  271  by way of cable of several kilometers in length. This is due to the following reason.  
           [0023]    It is known that any ultra-high speed electrical signal is unable to be transmitted over a very long distance by way of cable due to attenuation of the electrical signal. Taking this problem into consideration, the initial electrical signal is converted to an optical signal, the optical signal thus converted is transmitted through an optical transmission path, and then, the optical signal thus transmitted is reconverted to a resultant electrical signal on purpose. From this point of view, it is not meaningless to electrically interconnect the oscillator  262  with the oscillator  271  by way of cable of several kilometers in length. This means that this is not realistic for practical use.  
           [0024]    Thus, with the prior-art apparatus  200 , because of the necessity to electrically interconnect the two oscillators  271  and  262  with each other, it is unable to substantially synchronize the phase of the target light LT O  with the phase of the sampling light LT S  if the oscillators  271  and  262  are located far away from each other. As a result, there is a problem that the fluctuation of the time difference δt (i.e., the mutual jitter) is not decreased, thereby degrading the time resolution.  
           [0025]    Additionally, an apparatus and method for measuring the waveform of target light are disclosed in the Japanese Non-Examined Patent Publication No. 9-138165 published in 1997, in which sampling light with the repetition frequency that follows the fluctuation of repetition frequency of the target light is generated. In this apparatus and method, the nonlinear optical effects between the target light and the sampling light are used to measure the waveform of the target light and to control the repetition frequency of the sampling light.  
           [0026]    However, these apparatus and method of the Publication No. 9-138165 does not refer to the problem of the degradation of the time resolution in measurement.  
         SUMMARY OF THE INVENTION  
         [0027]    Accordingly, an object of the present invention is to provide a method and an apparatus for measuring she waveform of light that make it possible to synchronize easily the phase of sampling light with the phase of target light even if the target light is in the form of ultra-high speed pulses (e.g., 40 Gb/s or higher in repetition frequency) and is transmitted by way of a long transmission path (e.g., several kilometers in length).  
           [0028]    Another object of the present invention is to provide a method and an apparatus for measuring the waveform of light that make it possible to measure the waveform of target light with sufficient time resolution in real time even if the target light is in the form of ultra-high speed pulses (e.g., 40 Gb/s or higher in repetition frequency) and is transmitted by way of a long transmission path (e.g., several kilometers in length).  
           [0029]    The above objects together with others not specifically mentioned will become clear to those skilled in the art from the following description.  
           [0030]    According to a first aspect of the present invention, a method of measuring a pulse waveform of target light is provided, which comprises the steps of:  
           [0031]    (a) generating sampling light having a pulse width narrower than that of target light from the target light;  
           [0032]    a repetition frequency of the sampling light having a constant difference with respect to a repetition frequency of the target light;  
           [0033]    (b) supplying the sampling light and the target fight to a nonlinear optical member to generate cross-correlated light between the sampling light and the target light; and  
           [0034]    (c) measuring a waveform of the target light based on the cross-correlated light.  
           [0035]    With the method according to the first aspect of the present invention, in the step (a), the sampling light having a pulse width narrower than that of the target light is generated from the target light. In the step (b), the sampling light and the target light are supplied to the nonlinear optical member to generate the cross-correlated light between the sampling light and the target light. In the step (c), the waveform of the target light is measured based on the cross-correlated light. Moreover, the repetition frequency of the sampling light has the constant difference with respect to the repetition frequency of the target light.  
           [0036]    As a result, the phase of the sampling light can be easily synchronized with the phase of the target light even if the target light is ultra-high speed pulsed light and is transmitted by way of a long transmission channel. Thus, the waveform of the target light can be measured with sufficient time resolution in real time.  
           [0037]    In a preferred embodiment of the method according to the first aspect, the step (a) of generating the sampling light comprises the substeps of:  
           [0038]    (a- 1 ) extracting clock light from the target light;  
           [0039]    the clock light being synchronized with the target light;  
           [0040]    (a- 2 ) generating an electrical clock signal from the clock light;  
           [0041]    (a- 3 ) generating an electrical driving signal in such a way that a constant frequency difference exists between a frequency of the electrical driving signal and a frequency of the electrical clock signal; and  
           [0042]    (a- 4 ) generating the sampling light based on the electrical driving signal.  
           [0043]    In another preferred embodiment of the method according to the first aspect, in the substep (a- 1 ) of extracting the clock light from the target light, the target light is supplied to a passive mode-locked laser, thereby generating the clock light. The clock light has a repetition frequency (1/N) times as much as a repetition frequency of the target light, where N is a natural number.  
           [0044]    It is preferred in this preferred embodiment that a passive mode-locked semiconductor laser is used as the passive mode-locked laser.  
           [0045]    In this preferred embodiment of the method according to the first aspect, in the substep (a- 3 ) of generating the electrical driving signal,  
           [0046]    an electrical offset signal is generated by frequency-dividing the clock signal;  
           [0047]    an electrical difference-frequency signal is generated by frequency-mixing the electrical offset signal and the electrical difference-frequency signal together; and  
           [0048]    the electrical driving signal is generated based on a phase difference between the electrical offset signal and the electrical difference-frequency signal.  
           [0049]    Preferably, there may be additionally provided with the steps of:  
           [0050]    (d) generating an electrical cross-correlated signal from the cross-correlated light;  
           [0051]    (e) sampling peak values of the electrical cross-correlated signal; and  
           [0052]    (f) displaying an eye pattern corresponding to the electrical cross-correlated signal on a screen of a display device by repeatedly writing waveforms of the electrical cross-correlated signal while the electrical offset signal is used as a trigger.  
           [0053]    Preferably, the step (e) of sampling the peak values of the electrical cross-correlated signal is performed to be synchronized with an electrical signal generated by frequency-dividing the electrical driving signal.  
           [0054]    In still another preferred embodiment of the method according to the first aspect, in the substep (a- 3 ) of generating the electrical driving signal,  
           [0055]    an electrical offset signal is generated by frequency-dividing the clock signal;  
           [0056]    an electrical difference-frequency signal is generated by frequency-mixing the electrical offset signal and the electrical clock signal together; and  
           [0057]    the electrical driving signal is generated based on a phase difference between the electrical driving signal and the electrical difference-frequency signal.  
           [0058]    Preferably, there are additionally provided with the steps of:  
           [0059]    (d) generating an electrical cross-correlated signal from the cross-correlated light;  
           [0060]    (e) sampling peak values of the electrical cross-correlated signal; and  
           [0061]    (f) displaying an eye pattern corresponding to the electrical cross-correlated signal on a screen of a display device by repeatedly writing waveforms of the electrical cross-correlated signal while the electrical offset signal is used as a trigger.  
           [0062]    The step (e) of sampling the peak values of the electrical cross-correlated signal is preferably performed to be synchronized with an electrical signal generated by frequency-dividing the electrical driving signal.  
           [0063]    In a further preferred embodiment of the method according to the first aspect, in the substep (a- 3 ) of generating the electrical driving signal,  
           [0064]    an electrical offset signal is generated by frequency-dividing the clock signal;  
           [0065]    an electrical, additional driving signal is generated by frequency-dividing the electrical driving signal; and  
           [0066]    the electrical driving signal is generated based on a phase difference between the electrical offset signal and the electrical, additional driving signal.  
           [0067]    In this embodiment, preferably, there are additionally provided with the steps of:  
           [0068]    (d) generating an electrical cross-correlated signal from the cross-correlated light;  
           [0069]    (e) sampling peak values of the electrical cross-correlated signal; and  
           [0070]    (f) displaying an eye pattern corresponding to the electrical cross-correlated signal on a screen of a display device by repeatedly writing waveforms of the electrical cross-correlated signal while the electrical offset signal is used as a trigger.  
           [0071]    Preferably, the step (e) of sampling the peak values of the electrical cross-correlated signal is performed to be synchronized with an electrical signal generated by frequency-dividing the electrical driving signal.  
           [0072]    According to a second aspect of the present invention, an apparatus for measuring a pulse waveform of target light is provided, which comprises:  
           [0073]    (a) a sampling light generator for generating sampling light having a pulse width narrower than that of target light from the target light;  
           [0074]    a repetition frequency of the sampling light having a constant difference with respect to a repetition frequency of the target light;  
           [0075]    (b) a cross-correlated light generator for generating a cross-correlated light between the sampling light and the target light by supplying the sampling light and the target light to a nonlinear optical member; and  
           [0076]    (c) a measuring device for measuring a waveform of the target light based on the cross-correlated light.  
           [0077]    With the apparatus according to the second aspect of the present invention, because of the same reason as explained in the method according to the first aspect, the phase of the sampling light can be easily synchronized with the phase of the target light even if the target light is ultra-high speed pulsed light and is transmitted by way of a long transmission path. Thus, the waveform of the target light can be measured with sufficient time resolution in real time.  
           [0078]    In a preferred embodiment of the apparatus according to the second aspect, the sampling light generator (a) comprises:  
           [0079]    (a- 1 ) a clock light extractor for extracting clock light from the target light;  
           [0080]    the clock light being synchronized with the target light;  
           [0081]    (a- 2 ) an electrical clock signal generator for generating an electrical clock signal from the clock light;  
           [0082]    (a- 3 ) an electrical driving signal generator for generating an electrical driving signal in such a way that a constant frequency difference exists between a frequency of the electrical driving signal and a frequency of the electrical clock signal; and (a- 4 ) a sampling light generator for generating the sampling light based on the electrical driving signal.  
           [0083]    In this embodiment, preferably, the clock light extractor (a- 1 ) comprises a passive mode-locked laser;  
           [0084]    the target light being supplied to the passive mode-locked laser, thereby generating the clock light;  
           [0085]    the clock light having a repetition frequency (1/N) times as much as a repetition frequency of the target light, where N is a natural number.  
           [0086]    A passive mode-locked semiconductor laser is preferably used as the passive mode-locked laser.  
           [0087]    In another preferred embodiment of the apparatus according to the second aspect, the electrical driving signal generator (a- 3 ) comprises:  
           [0088]    an electrical offset signal generator for generating an electrical offset signal by frequency-dividing the clock signal; and  
           [0089]    an electrical difference-frequency signal generator for generating an electrical difference-frequency signal by frequency-mixing the electrical offset signal and the electrical difference-frequency signal together;  
           [0090]    wherein the electrical driving signal is generated based on a phase difference between the electrical offset signal and electrical difference-frequency signal.  
           [0091]    In this embodiment, preferably, there are additionally provided with  
           [0092]    (d) an electrical cross-correlated signal generator for generating an electrical cross-correlated signal from the cross-correlated light;  
           [0093]    (e) a peak value sampler for sampling peak values of the electrical cross-correlated signal; and  
           [0094]    (f) a display device for displaying an eye pattern corresponding to the electrical cross-correlated signal on a screen of a display device by repeatedly writing waveforms of the electrical cross-correlated signal while the electrical offset signal is used as a trigger.  
           [0095]    Preferably, the peak value sampler (e) samples the peak values of the electrical cross-correlated signal to be synchronized with an electrical signal generated by frequency-dividing the electrical driving signal.  
           [0096]    In still another preferred embodiment of the apparatus according to the second aspect, in the electrical driving signal generator (a- 3 );  
           [0097]    an electrical offset signal  4  is generated by frequency-dividing the clock signal;  
           [0098]    an electrical difference-frequency signal is generated by frequency-mixing the electrical offset signal and the electrical clock signal together; and  
           [0099]    the electrical driving signal is generated based on a phase difference between the electrical driving signal and electrical difference-frequency signal.  
           [0100]    In this embodiment, preferably, there are additionally provided with  
           [0101]    (d) an electrical cross-correlated signal generator for generating an electrical cross-correlated signal from the cross-correlated light;  
           [0102]    (e) a peak value sampler for sampling peak values of the electrical cross-correlated signal; and  
           [0103]    (f) a display device for displaying an eye pattern corresponding to the electrical cross-correlated signal on a screen of a display device by repeatedly writing waveforms of the electrical cross-correlated signal while the electrical offset signal is used as a trigger.  
           [0104]    Preferably, the peak value sampler operates to be synchronized with an electrical signal generated by frequency-dividing the electrical driving signal.  
           [0105]    In a further preferred embodiment of the apparatus according to the second aspect, in the electrical driving signal generator (a- 3 );  
           [0106]    an electrical offset signal is generated by frequency-dividing the clock signal;  
           [0107]    an electrical, additional driving signal is generated by frequency-dividing the electrical driving signal; and  
           [0108]    the electrical driving signal is generated based on a phase difference between the electrical offset signal and the electrical, additional driving signal.  
           [0109]    In this embodiment, preferably, there are additionally provided with  
           [0110]    (d) an electrical cross-correlated signal generator for generating an electrical cross-correlated signal from the cross-correlated light;  
           [0111]    (e) a peak value sampler for sampling peak values of the electrical cross-correlated signal; and  
           [0112]    (f) a display device for displaying an eye pattern corresponding to the electrical cross-correlated signal on a screen of a display device by repeatedly writing waveforms of the electrical cross-correlated signal while the electrical offset signal is used as a trigger.  
           [0113]    Preferably, the peak value sampler operates to be synchronized with an electrical signal generated by frequency-dividing the electrical driving signal. 
       
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS  
       [0114]    In order that the present invention may be readily carried into effect, it will now be described with reference to the accompanying drawings.  
         [0115]    [0115]FIG. 1 is a schematic block diagram showing the configuration of a prior-art optical measuring apparatus  
         [0116]    [0116]FIG. 2 is a schematic block diagram showing the configuration of an optical measuring apparatus according to a first embodiment of the invention.  
         [0117]    [0117]FIG. 3 is a schematic illustration showing the configuration of the optical clock extractor used in the apparatus according to the first embodiment of FIG. 2.  
         [0118]    [0118]FIG. 4 is a schematic block diagram showing the configuration of the PLL circuit used in the apparatus according to the first embodiment of FIG. 2.  
         [0119]    [0119]FIG. 5 is a schematic block diagram showing the configuration of an optical measuring apparatus according to a second embodiment of the invention.  
         [0120]    [0120]FIG. 6 is a schematic block diagram showing the configuration of an optical measuring apparatus according to a third embodiment of the invention.  
         [0121]    [0121]FIG. 7 is a schematic block diagram showing the configuration of an optical measuring apparatus according to a fourth embodiment of the invention.  
         [0122]    [0122]FIG. 8 is a schematic block diagram showing the configuration of an optical measuring apparatus according to a fifth embodiment of the invention.  
         [0123]    [0123]FIG. 9 is a schematic block diagram showing the configuration of an optical measuring apparatus according to a sixth embodiment of the invention.  
         [0124]    [0124]FIG. 10 is a schematic block diagram showing the configuration of an optical measuring apparatus according to a seventh embodiment of the invention.  
         [0125]    [0125]FIG. 11 is a schematic block diagram showing the configuration of an optical measuring apparatus according to an eighth embodiment of the invention.  
         [0126]    [0126]FIG. 12 is a schematic block diagram showing the configuration of an optical measuring apparatus according to a ninth embodiment of the invention.  
         [0127]    [0127]FIG. 13 is a schematic block diagram showing the configuration of an optical measuring apparatus according to a tenth embodiment of the invention.  
         [0128]    [0128]FIG. 14 is a waveform diagram showing the waveforms of the target light, the sampling light, the cross-correlated light, and the eye pattern. 
     
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0129]    Preferred embodiments of the present invention will be described in detail below while referring to the drawings attached.  
       FIRST EMBODIMENT  
       [0130]    An apparatus for measuring optically the waveform of sampling light according to a first embodiment of the invention has the structure shown in FIGS. 2, 3, and  4 , in which thick lines indicate the flow of optical signals while thin lines indicate the flow of electrical signals  
         [0131]    The optical measuring apparatus  100  according to the first embodiment of the invention comprises an optical clock extractor  1 , a photodetector  2 , a band-pass filter (BPF)  3 , a phase-locked loop (PLL) circuit  4 , a sampling light source  5 , a nonlinear optical member  6 , an optical filter  7 , a photodetector  8 , an electrical signal processing circuit  9 , and a display device  10 .  
         [0132]    The optical clock extractor  1  receives part of pulsed target light LT O  with a repetition frequency of f O , the target light LT O  has been transmitted from an external apparatus (not shown) by way of a long-distance optical path (e.g., optical fibers). The extractor  1  generates clock light LT CLK  with a repetition frequency of (f O /N) synchronized with the target light LT O  thus received. The clock light LT CLK  is then applied to the photodetector  2 .  
         [0133]    An example of the configuration of the optical clock extractor  1  is shown in FIG. 3. In this configuration, the extractor  1  comprises an optical isolator  11 , a lens  12 , a passive mode-locked semiconductor laser  13 , a biasing voltage source  14 , a direct current (DC) current source  15 , a lens  16 , and an optical isolator  17 .  
         [0134]    With the laser  13 , it is supposed that the resonator length is L, the refractive index containing the group velocity dispersion is n, the repetition frequency in the passive mode-locked operation is f M , and the velocity of light in vacuum is c. In this case, the repetition frequency f M  is expressed by the following equation (2).  
               f   M     ≅     c     2      nL               (   2   )                               
 
         [0135]    To cause the laser  13  to perform the passive mode-locked operation, the resonator length L is adjusted to satisfy the following equation (3), where N is a natural number (i.e., N=1, 2, 3, 4, . . . ). At the same time as this, a reverse bias voltage V B  is applied to the suturable absorption region  13   a  of the laser  13  by the voltage source  14  and a forward DC current I f  is supplied to the gain region  13   b  of the laser  13  by the current source  15 .  
               f   M     ≅       f   0     N             (   3   )                               
 
         [0136]    When the target light LT O  with the repetition frequency f O is applied to the laser  13  in the passive mode-locked operation by way of the isolator  11  and the lens  12 , the light LT O  is modulated in the suturable absorption region  13   a  of the laser  13 , thereby converting the repetition frequency f O  to the repetition frequency (f O /N). Thus, the clock light LT CLK  with the repetition frequency (f O /N), which is synchronized with the target light LT O , is generated in the laser  13 . The clock light LT CLK  thus generated in the laser  13  is emitted from the extractor  1  by way of the lens  16  and the isolator  17 .  
         [0137]    The possible increase of jitter in the clock light LT CLK  in the extractor  1  can be limited to the order of femtoseconds (fs) by suitably adjusting the operation condition of the extractor  1 . Thus, the increase of jitter can be ignored in the measurement.  
         [0138]    Referring to FIG. 2, the photodetector  2  receives the clock light LT CLK  emitted from the optical clock extractor  1  and photo electrically converts the light LT CLK  thus received to an electrical clock signal S CLK . The clock signal S CLK  is then sent to the BPF  3 .  
         [0139]    The BPF  3  has a characteristic that allows only the electrical signals with the frequency in the vicinity of (f O /N) to pass through the same. Thus, only the electrical clock signal S CLK  with the frequency (f O /N) passes through the BPF  3 , resulting in an electrical sinusoidal signal S SIN  with the frequency (f O /N). The signal S SIN  is then sent to the PLL circuit  4 .  
         [0140]    The PLL circuit  4  receives the electrical sinusoidal signal S SIN  with the frequency (f O /N) and generates an electrical driving signal S D  and an electrical offset signal S OST . The driving signal S D  has a frequency f S , which is different from the frequency (f O /N) of the sinusoidal signal S SIN  by a constant frequency difference Δf. The offset signal S OST  has a frequency equal to the frequency difference Δf. The circuit  4  sends the driving signal S D  and the offset signal S OST  to the signal processing circuit  9  and the display device  10 , respectively.  
         [0141]    An example of the configuration of the PLL circuit  4  is shown in FIG. 4, which comprises a frequency mixer  21 , a low-pass filter (LPF)  22 , a frequency divider  23 , a phase comparator  24 , and a voltage-controlled oscillator (VCO)  25 .  
         [0142]    The frequency mixer  21  receives the sinusoidal signal S SIN  with the frequency (f O /N) supplied from the BPF  3  and the driving signal S D  with the frequency f S  [=(f O /N)−Δf] supplied from the VCO  25  and then, frequency-mixes them together. Thus, the mixer  21  generates an electrical mix signal S MIX  and outputs the same. The mix signal S MIX  contains a difference-frequency component with a difference-frequency of [(f 0 /N)−f S ] and a sum-frequency component with a sum-frequency of [(f O /N)+f S ] and their harmonics.  
         [0143]    The LPF  22  allows only the difference-frequency component with the difference-frequency [(f O /N)−f S ] and therefore, it outputs the same component as a difference-frequency signal S DIF  with the difference-frequency [(f O /N)−f S ] to the phase comparator  24 .  
         [0144]    The frequency divider  23  receives the sinusoidal signal S SIN  with the frequency (f O /N) supplied from the BPF  3  and then, frequency-divides the signal S SIN  by a dividing factor M, where M is a natural number (i.e. , M=1, 2, 3, 4, . . . ). Thus, the divider  23  outputs the electrical offset signal S OST  with the frequency Δf to the phase comparator  24  and the display device  10 , where Δf satisfies the relationship of Δf=[f O /(N·M)].  
         [0145]    The frequency divider  23  is, for example, formed by a digital logic circuit such as an emitter-coupled logic (ECL) capable of ultra-high-speed operation. In this case, the rise time of the offset signal S OST  is sufficiently short and the fluctuation thereof is sufficiently small. Thus, there is an advantage that the eye pattern is displayed clearly in the display device  10 , raising the time resolution in the measuring apparatus  100  according to the first embodiment of FIG. 2.  
         [0146]    The phase comparator  24  compares the phase of the difference-frequency signal S DIF  with the difference-frequency [(f O /N)−f S ] and that of the offset signal S OST  with the frequency Δf, outputting an error signal S ER  to the VCO  25 . The error signal S ER  thus outputted is proportional to the phase difference between these two signals S DIF  and S OST .  
         [0147]    The VCO  25  generates the electrical driving signal S D  whose frequency is kept at f S  according to the voltage (i.e., amplitude) of the error signal S ER  from the phase comparator  24 , where f S =(f O /N)−Δf. Then, the VCO  25  outputs the driving signal S D  to the frequency mixer  21 , the signal processing circuit  9  and the sampling light source  5 .  
         [0148]    Here, the explanation is returned to the apparatus  100  itself again. As shown in FIG. 2, the sampling light source  5  emits the sampling light LT S  in the form of narrow pulses according to the electrical driving signal S D  supplied from the PLL circuit  4 . The sampling light LT S  has a repetition frequency f S [=(f O /N)−Δf]. This means that the repetition frequency f S  of the sampling light LT S  is different from the frequency (f O /N) of the clock light LT CLK  by Δf, where f O  is the repetition frequency of the target light LT O .  
         [0149]    As the sampling light source  5 , for example, any mode-locked laser with the Fabry-Perot resonator (eg., a mode-locked semiconductor laser), any mode-locked laser with the ring resonator (eg., an optical fiber laser), or any gain-switching semiconductor laser may be used. The details of the lasers of this type are, for example, disclosed in the above-referred Japanese Non-Examined Patent Publication Nos. 8-29814 and 9-160082.  
         [0150]    The nonlinear optical member  6  is made of a ferroelectric crystal such as KTP that causes a secondary nonlinear optical effect (e.g., SFG), or a semiconductor optical amplifier or a quartz-system optical waveguide such as an optical fiber that causes a tertiary nonlinear optical effect (e.g., FWM). The member  6  emits intensity cross-correlated light LT CC  between the pulses of the target light LT O  supplied from the external apparatus and the sampling light LT S  supplied from the sampling light source  5 . The intensity cross-correlated light LT CC  thus emitted has a repetition frequency of f S , which is equal to the frequency of the electrical driving signal S D .  
         [0151]    Background light LT B  is emitted from the optical member  6  along with the cross-correlated light LT CC . The background light LT B  contains the target light LT O  and the sampling light LT S  and their secondary or higher harmonics.  
         [0152]    Here, the optical frequencies of the target light LT O  and the sampling light LT S  are defined as ν sig  and ν smp , respectively. In this case, if the optical member  6  is made of a material causing a secondary nonlinear optical effect, sum-frequency light with the optical frequency ν sum  given by the following equation (4) or difference-frequency light with the optical frequency ν dif  given by the following equation (5) is generated in the member  6 . The sum-frequency or difference-frequency light thus generated is outputted from the member  6  as the cross-correlated light LT CC .  
         ν sum =ν sig  +ν smp    (4)  
         ν dif =|ν sig −ν smp |  (5)  
         [0153]    On the other hand, if the optical member  6  is made of a material causing a tertiary nonlinear optical effect, four-wave mixed light with the optical frequency ν fwm  given by the following equation (6) is generated in the member  6 . The four-wave mixed light thus generated is outputted from the member  6  as the cross-correlated light LT CC .  
         ν fwm =2ν smp −ν sig    (6)  
         [0154]    The optical filter  7  removes the background light LT B  of the cross-correlated light LT CC  emitted from the optical member  6 , allowing only the cross-correlated light LT CC  to pass through the same. The cross-correlated light LT CC  is then sent to the photodetector  8 .  
         [0155]    The photodetectcr  8  converts photoelectrically the cross-correlated light LT CC  to an electrical pulsed signal S CC  and then, sends the signal S CC  thus generated to the signal processing circuit  9 . As described previously in the explanation about the prior-art apparatus  200 , the photodetector  8  needs to have a frequency band equal to or higher than the repetition frequency f S  the sampling light LT S .  
         [0156]    The signal processing circuit  9  samples or extracts the peak values of the electrical pulsed signal S CC  supplied from the photodetector  8 . The sampling operation of the circuit  9  is conducted to be synchronized with the driving signal S D  with the repetition frequency f S  equal to the sampling frequency of the sampling light LT S . The peak values thus extracted are plotted with the circular dots in FIG. 14 in the vicinity of the waveform c. The circuit  9  generates an electrical peak signal S CCP  corresponding to the extracted peak values of the signal S CC , supplying the peak signal S CCP  to the display device  10 .  
         [0157]    The display device  10  displays an eye pattern as shown by the waveform d in FIG. 14 on its screen (not shown) by repeatedly writing a plurality of waveforms on the same plane at a specific period of [1/(N·Δf)] on the basis of the peak signal S CCP  supplied from the signal processing circuit  9 . The displaying operation of the device  10  is carried out while the offset signal S OST  with the frequency Δf supplied by the PLL circuit  4  is used as a trigger signal.  
         [0158]    Next, the operation of the measuring apparatus  100  according to the first embodiment having he above-described configuration is explained below.  
         [0159]    First, the optical clock extractor  1  receives the pulsed target light LT O  with the repetition frequency f O  and emits the pulsed clock light LT CLK  with the repetition frequency (f O /N) synchronized with the target light LT O  thus received. The target light LT O  has the waveform a shown in FIG. 14. The waveform of the clock light LT CLK  is not shown in FIG. 14; however, the waveform of the clock light LT CLK  is approximately the same as the waveform b of the sampling light LT S  shown in FIG. 14. The repetition frequency (f O /N) of the clock light LT CLK  is different from that of the sampling light LT S  by Δf.  
         [0160]    The clock light LT CLK  thus emitted from the extractor  1  is photoelectrically converted to the electrical clock signal S CLK  by the photodetector  2 . Only the electrical clock signal S CLK  with the frequency (f O /N) can pass through the BPF  3  and as a result, the electrical sinusoidal signal S SIN  with the frequency of (f O /N) is outputted from the BPF  3  and supplied to the PLL circuit  4 .  
         [0161]    In the PLL circuit  4 , as shown in FIG. 4, the sinusoidal signal S SIN  with the frequency (f O /N) is frequency-divided by the dividing factor M by the frequency divider  23 , generating the electrical offset signal S OST  with the frequency Δf. The offset signal S OST  is then supplied to the phase comparator  24  and at the same time, is supplied to the display device  10  as the trigger signal.  
         [0162]    On the other hand, the sinusoidal signal S SIN  with the frequency (f O /N) is supplied to the frequency mixer  21  also. In the mixer  21 , the sinusoidal signal S SIN  is mixed with the driving signal S D  with the frequency f S  supplied by the VCO  25 . Thus, the electrical mix signal S MIX  is generated. Only the difference-frequency component with the difference-frequency of [(f O /N)−f S ] of the mix signal S MIX  is selectively outputted by the LPF  22  as the difference-frequency signal S DIF  with the difference-frequency [(f O /N)−f S ].  
         [0163]    The difference-frequency signal S DIF  and the offset signal S OST  are sent to the phase comparator  24  and compared in phase with each other. Thus, the error signal S ER  proportional to the phase difference between these two signals S DIF  and S OST  is outputted from the comparator  24 . The error signal S ER  thus outputted is then supplied to the VCO  25 .  
         [0164]    In the VCO  25 , the driving signal S D  whose frequency is kept at f S  according to the voltage of the error signal S ER  is generated, where f S =(f 0 /N)−Δf. Then, the driving signal S D  thus generated is hen sent to the frequency mixer  21 , the signal processing circuit  9 , and the sampling light source  5 .  
         [0165]    The electrical offset signal S OST  (frequency: Δf=f O /(N·M)) outputted from the PLL circuit  4  is generated by frequency-dividing the electrical sinusoidal signal S SIN  (frequency: f O /N) generated from the clock light LT CLK  (repetition frequency: f O /N), where the clock light LT CLK  is synchronized with the target light LT O  (repetition frequency: f O ). Therefore, the offset signal S OST  is synchronized with the target light LT O . In this case, if the frequency of the driving signal S D  outputted from the VCO  25  is kept at f S  , the phase of the driving signal S D  is always synchronized with the phase of the target light LT 0 .  
         [0166]    The driving signal S D  thus generated is supplied to the sampling light source  5 . On the other hand, as explained previously, the driving signal S D  has the frequency f S  and the phase synchronized with the phase of the target light LT O . Accordingly, the sampling light LT S  emitted from the light source  5  is in the form of narrow pulses with the repetition frequency f S , as shown by the waveform b in FIG. 14.  
         [0167]    The sampling light LT S  thus emitted from the source  5  is supplied to the nonlinear optical member  6  while the target light LT O  is supplied to the same member  6  as well. Thus, the cross-correlated light LT CC  between the sampling light LT S  and the target light LT O  is emitted from the member  6  along with the background light LT B . The cross-correlated light LT CC  has the waveform c shown in FIG. 14. The cross-correlated light LT CC  and the background light LT B  enter the optical filter  7 . Only the cross-correlated light LT CC  passes through the filter  7  and enters the photodetector  8 . In the photodetector  8 , the cross-correlated light LT CC  is photoelectrically converted to the electrical pulsed signal S CC  and then, it is emitted from photodetector  8 .  
         [0168]    The electrical signal S CC  from the photodetector  8  is subjected to the specific signal processing operation in the electrical signal processing circuit  9 . Then, the peak values (i.e., the circular dots in FIG. 14) of the electrical signal S CC  are sampled and then, supplied to the display device  10  as the electrical peak signal S CCP .  
         [0169]    In the display device  10 , an eye pattern is displayed on the screen as shown by the waveform d in FIG. 14. This is realized by repeatedly writing a plurality of waveforms on the same plane at a specific period of [1/(N·Δf)] on the basis of the peak signal S CCP  supplied from the signal processing circuit  9 . The offset signal S OST  with the frequency Δf supplied by the PLL circuit  4  is used as a trigger signal in the device  10 .  
         [0170]    A measuring person or tester evaluates the characteristics of a target optical transmission system to be evaluated according to the opening level of the eye pattern d displayed on the screen of the display device  10 .  
         [0171]    With the optical measuring device  100  according to the first embodiment, as explained in detail, the electrical driving signal S D  (frequency: f S ) synchronized in phase with the target light LT O  (repetition frequency: f O ) is generated from the target light LT O  itself in the PLL circuit  4 . Also, the sampling light LT S  (repetition frequency: f S ) is generated in the sampling light source  5  on the basis of the electrical driving signal S D  thus generated. Thus, unlike the prior-art apparatus  200  shown in FIG. 1, the electrical connection between the driving signal oscillator  262  and the driving signal oscillator  271  provided in the external apparatus  261  is unnecessary.  
         [0172]    As a result, even when the ultra-high speed pulses of the target light LT O  having a repetition frequency that exceeds 40 Gb/s, which has been transmitted by way of an optical transmission path (which is formed by, e.g., optical fibers) of several kilometers in length, is measured in real time, the target light LT O  and the sampling light LT S  can be synchronized in phase with each other. Accordingly, the fluctuation of the time difference δt (i.e., the mutual jitter) can be decreased, which raises the time resolution in measurement.  
       SECOND EMBODIMENT  
       [0173]    [0173]FIG. 5 shows a PLL circuit  31  provided in an apparatus for measuring the waveform of sampling light according to a second embodiment of the invention.  
         [0174]    The apparatus according to the second embodiment has a configuration obtained by replacing the PLL circuit  4  in the apparatus  100  according to the first embodiment with the a circuit  31 . The other configuration of the apparatus of the second embodiment is the same as the apparatus  100  of the first embodiment. Thus the explanation about only the PLL circuit  31  is presented here for the sake of simplification of description.  
         [0175]    As shown in FIG. 5, the PLL circuit  31  comprises a frequency divider  32 , a frequency mixer  33 , a BPF  34 , a phase comparator  35 , and a VCO  36 .  
         [0176]    The frequency divider  32  receives the electrical sinusoidal signal S SIN  with the frequency (f O /N) supplied from the BPF  3  and then, frequency-divides the signal S SIN  by the dividing factor M, where M is a natural number. Thus, the divider  32  outputs the electrical offset signal S OST  with the frequency Δf to the frequency mixer  33  and the display device  10 , where Δf satisfies the relationship of Δf=[f O /(N·M)].  
         [0177]    The frequency divider  32  is, for example, formed by a digital logic circuit such as ECL capable of ultra-high-speed operation.  
         [0178]    The frequency mixer  33  receives the sinusoidal signal S SIN  with the frequency (f O /N) supplied from the BPF  3  and the electrical offset signal S OST  with the frequency Δf supplied from the divider  32  and then, frequency-mixes them together. Thus, the mixer  33  generates the electrical mix signal S MIX  and outputs the same to the BPF  34 . The mix signal S MIX  contains a difference-frequency component with a difference-frequency [(f O /N)−Δf] and a sum-frequency component with a sum-frequency [(f O /N)+Δf] and their harmonics.  
         [0179]    The BPF  34  allows only the difference-frequency component with the difference-frequency [(f O /N)−Δf] and outputs the same component as the electrical difference-frequency signal S DIF  with the difference-frequency [(f O /N)−Δf] to the phase comparator  35 .  
         [0180]    The phase comparator  35  compares the phase of the difference-frequency signal S DIF  with the difference-frequency [(f O /N)−Δf] and that of the driving signal S D  with the frequency f S  [=(f O /N)−Δf] supplied from the VCO  36 . Thus, the comparator  35  outputs an electrical error signal S ER  to the VCO  36 . The error signal S ER  is proportional to the phase difference between these two signals S DIF  and S D .  
         [0181]    The VCO  36  generates the driving signal S D  whose frequency is kept at f S [=(f O /N)−Δf] according to the voltage (i.e., the amplitude) of the error signal S ER  from the phase comparator  35 . Then, the VCO  36  outputs the driving signal S D  to the phase comparator  35  and the signal processing circuit  9 . Also, the driving signal S D  is fed back to the VCO  36 .  
         [0182]    Next, the operation of the PLL circuit  31  of FIG. 5 having the above-described configuration is explained below.  
         [0183]    In the PLL circuit  31 , the electrical sinusoidal signal S SIN  with the frequency of (f O /N) is frequency-divided by the dividing factor M by he frequency divider  32 , generating the offset signal S OST  with the frequency Δf. The offset signal S OST  is supplied to the frequency mixer  33  and the display device  10  as the trigger signal.  
         [0184]    On the other hand, the sinusoidal signal S SIN  with the frequency of (f O /N) is frequency-mixed with the offset signal S OST  with the frequency Δf in the frequency mixer  33 , thereby outputting the mix signal S MIX  to the BPF  34 . The BPF  34  allows only the difference-frequency component of the mixer signal S MIX  to pass through the same, outputting the component as the difference-frequency signal S DIF  with the difference-frequency [(f O /N)−Δf].  
         [0185]    In the phase comparator  35 , the error signal S ER  proportional to the phase difference between the difference signal S DIF  and the driving signal S D  is generated and sent to the VCO  36 . In the VCO  36 , the frequency of the driving signal S D  is kept at f S  according to the voltage of the error signal S ER .  
         [0186]    With the measuring apparatus according to the second embodiment comprising the PLL circuit  31  in FIG. 5, the electrical offset signal S OST  (frequency: Δf=f O /(N·M)) outputted from the frequency divider  32  is generated by frequency-dividing the electrical sinusoidal signal S SIN  (frequency: f O /N) generated from the clock light LT CLK  (repetition frequency: f O /N), where the clock light LT CLK  is synchronized with the target light LT O  (repetition frequency: f O ) Therefore, the offset signal S OST  is synchronized with the target light LT O . In this case, since the frequency of the driving signal S D  outputted from the VCO  36  is kept at f S , the phase of the driving signal S D  is always synchronized with the phase of the target light LT O .  
         [0187]    As a result, there are approximately the same advantages as those in the measuring device  100  according to the first embodiment.  
       THIRD EMBODIMENT  
       [0188]    [0188]FIG. 6 shows a PLL circuit  41  provided in an apparatus for measuring the waveform of sampling light according to a third embodiment of the invention.  
         [0189]    The apparatus according to the third embodiment has a configuration obtained by replacing the PLL circuit  4  in the apparatus  100  according to the first embodiment with the PLL circuit  41 . The other configuration of the apparatus of the third embodiment is the same as the apparatus  100  of the first embodiment. Thus, the explanation about only the PLL circuit  41  is presented here for the sake of simplification of description.  
         [0190]    As shown in FIG. 6, the PLL circuit  41  comprises first and second frequency dividers  42  and  43 , a phase comparator  44 , and a VCO  45 .  
         [0191]    The first frequency divider  42  receives the sinusoidal signal S SIN  with the frequency (f O /N) supplied from the BPF  3  and the frequency-divides the signal S SIN  by a dividing factor M, where M is a natural number. Thus, the divider  42  outputs the electrical offset signal S OST  with the frequency Δf to the phase comparator  44  and the display device  10 , where Δf satisfies the relationship of Δf=[f O /(N·M)].  
         [0192]    The second frequency divider  43  receives the driving signal S D  with the frequency f S  [=(f O /N)−Δf] supplied from the VCO  45  and then, frequency-divides the signal S D  by a dividing factor K, where K is a natural number (i.e., K=1, 2, 3, 4, . . .). Thus, the divider  43  outputs an electrical second or additional driving signal S DD  with the frequency (f S /K) to the phase comparator  44 .  
         [0193]    The dividing factors M and K are determined in such a way that the frequency Δf [=f O /(N·M)] is equal to the frequency (f S /K).  
         [0194]    Each of the first and second frequency dividers  42  and  43  is, for example, formed by a digital logic circuit such as ECL capable of ultra-high-speed operation.  
         [0195]    The phase comparator  44  compares the phase of the offset signal S OST  with the frequency Δf[=f O /(N·M)] from the first frequency divider  42  and that of the second or additional driving signal S DD  with the frequency (f S /K) supplied from the second frequency divider  43 . Thus, the comparator  44  outputs an error signal S ER  to the VCO  45 . The error signal S ER  is proportional to the phase difference between these two signals S OST  and S DD  .  
         [0196]    The VCO  45  generates the driving signal S D  whose frequency is kept at f S  [=(f O /N)−Δf] according to the voltage of the error signal S ER  from the phase comparator  44 . Then, the VCO  45  outputs the driving signal S D  thus generated to the frequency divider  43 , the signal processing circuit  9 , and the sampling light source  5 .  
         [0197]    Next, the operation of the PLL circuit  41  having the above-described configuration is explained below.  
         [0198]    In the PLL circuit  41 , the electrical sinusoidal signal S SIN  with the frequency of (f O /N) is frequency-divided by the dividing factor M by the first frequency divider  42 , generating the offset signal S OST  with the frequency Δf. The offset signal S OST  is supplied to the phase comparator  44  and the display device  10  as the trigger signal.  
         [0199]    On the other hand, the driving signal S D  with the frequency f S supplied from the VCO  45  is frequency-divided by the dividing factor K by the second frequency divider  43 , generating the second/additional driving signal S DD  with the frequency (f S /K).  
         [0200]    The phase comparator  44  compares in phase the offset signal S OST  and the second/additional driving signal S DD , outputting the error signal S ER  proportional to the phase difference between them to the VCO  45 . The VCO  45  operates to keep the frequency of the driving signal S D  at f S  according to the voltage of the error signal S ER .  
         [0201]    With the apparatus according to the third embodiment comprising the PLL circuit  41  of FIG. 6, the electrical offset signal S OST  (frequency: Δf=f O /(N·M)) outputted from the first frequency divider  42  is generated by frequency-dividing the electrical sinusoidal signal S SIN  (frequency: f O /N) generated from the clock light LT CLK  (repetition frequency: f O /N), where the clock light LT CLK  is synchronized with the target light LT O  (repetition frequency: f O ) . Therefore, the offset signal S OST  is synchronized with the target light LT O . In this case, if the frequency of the driving signal S D  outputted from the VCO  45  is kept at f S  [=(f O /N)−Δf], the phase of the driving signal S D  is always synchronized with the phase of the target light LT O .  
         [0202]    As a result, there are approximately the same advantages as those in the measuring device  100  according to the first embodiment. Moreover, since the frequency mixer  21 , the LPF  22  or BPF  34  are unnecessary, the configuration is simpler than the first or second embodiment.  
       FOURTH EMBODIMENT  
       [0203]    [0203]FIG. 7 shows an apparatus  100 A for measuring the waveform of sampling light according to a fourth embodiment of the invention, which has the same configuration as the apparatus  100  according to the first embodiment of FIG. 2 except that a dispersion compensator  51  is additionally provided between the sampling light source  5  and the nonlinear optical member  6 . Therefore, the explanation about the same configuration and operation is omitted here for the simplification by attaching the same reference symbols as used in the first embodiment in FIG. 7.  
         [0204]    With the above-described apparatus  100  according to the first embodiment of FIG. 2, the sampling light LT S  emitted from the sampling light source  5  is directly supplied to the nonlinear optical member  6 . In this case, however, if the pulse width (on the time axis) of the sampling light LT S  is comparatively wide, there is a possibility that the time resolution of the apparatus  100  degrades.  
         [0205]    On the other hand, with the apparatus  100 A according to the fourth embodiment of FIG. 7, the sampling light LT S  emitted from the sampling light source  5  enters the nonlinear optical member  6  by way of the dispersion compensator  51 . Thus, the sampling light LT SDC  passing through the compensator  51  has the narrowed pulse widths. This eliminates the possibility that the time resolution of the apparatus  100 A degrades and at the same time, it may raise the time resolution.  
         [0206]    As the dispersion compensator  51 , any type of dispersion compensator may be used. Preferably, a dispersion-compensating optical fiber is used as the compensator  51 .  
       FIFTH EMBODIMENT  
       [0207]    [0207]FIG. 8 shows an apparatus  100 B for measuring the waveform or sampling light according to a fifth embodiment of FIG. 2, which has the same configuration as the apparatus  100  according to the first embodiment except that two polarization controllers  52   a  and  52   b  are additionally provided.  
         [0208]    The first polarization controller  52   a  for controlling the polarization state of incident light is provided between the input port or inlet (not shown) of the apparatus  100 B and the nonlinear optical member  6 . The second polarization controller  52   b  for controlling the polarization state of incident light is provided between the sampling light source  5  and the same member  6 . Therefore, the explanation about the same configuration is omitted here for the simplification by attaching the same reference symbols as used in the first embodiment in FIG. 8.  
         [0209]    With the above-described apparatus  100  according to the first embodiment of FIG. 2, the target light LT O  is directly supplied to the nonlinear optical member  6  while the sampling light LT S  emitted from the sampling light source  5  is directly supplied to the same member  6 .  
         [0210]    Unlike this, with the apparatus l 00 B according to the fifth embodiment of FIG. 8, the target light LT O  is supplied to the nonlinear optical member  6  by way of the first polarization controller  52   a  and at the same time, the sampling light LT S  emitted from the sampling light source  5  is supplied to the same member  6  by way of the second polarization controller  52   b . This is to optimize the polarization states of the target light LT O  and the sampling light LT S  prior to entering the member  6 , thereby increasing the intensity of the cross-correlated light LT CC  generated by the nonlinear optical effect in the member  6 . This is on the basis of the fact that the intensity of the cross-correlated light LT CC  varies dependent strongly on the polarization state of the incoming light LT O  and LT S .  
         [0211]    As a result, the apparatus  100 B according to the fifth embodiment has an additional advantage that the measurement sensitivity is raised compared with the apparatus  100  according to the first embodiment.  
       SIXTH EMBODIMENT  
       [0212]    [0212]FIG. 9 shows an apparatus  100 C for measuring the waveform of sampling light according to a sixth embodiment of the invention, which has the same configuration as the apparatus  100 B according to the fifth embodiment of FIG. 8 except that an optical coupler  53  is additionally provided.  
         [0213]    The coupler  53  is provided so as to couple the polarization-controlled target light LT OPC  outputted from the first polarization controller  52   a  with the polarization-controlled sampling light LT SPC  outputted from the second polarization controller  52   b . The polarization-controlled target light LT OPC  and the polarization-controlled sampling light LT SPC  thus coupled is supplied to the nonlinear optical member  6 . Therefore, the explanation about the same configuration is omitted here for the simplification by attaching the same reference symbols as used in the fifth embodiment in FIG. 9.  
         [0214]    With the drive  100 C according to the sixth embodiment of FIG. 9, the same advantages as those in the fifth embodiment are given.  
       SEVENTH EMBODIMENT  
       [0215]    [0215]FIG. 10 shows an apparatus  100 D for measuring the waveform of sampling light according to a seventh embodiment of the invention, which has the same configuration as the apparatus  100  according to the first embodiment of FIG. 2 except that two optical amplifiers  54   a  and  54   b  are additionally provided.  
         [0216]    The first optical amplifier  54   a  is provided between the input port (not shown) of the apparatus  100 D and the nonlinear optical member  6 . The second optical amplifier  54   b  is provided between the sampling light source  5  and the same member  6 . Therefore, the explanation about the same configuration is omitted here for the simplification by attaching the same reference symbols as used in the first embodiment in FIG. 10.  
         [0217]    With the above-described apparatus  100  according to the first embodiment, the target light LT O  and the sampling light LT S  is directly supplied to the nonlinear optical member  6 . Un like this, with the apparatus  100 D according to the seventh embodiment of FIG. 10, the amplified target light LT OA  outputted from the first optical amplifier  54   a  and the amplified sampling light LT S  outputted from the second optical amplifier  54   b  is supplied to the nonlinear optical member  6 . This is to enhance the intensity of the target light LT O  and the sampling light LT S  prior to entering the member  6 , thereby increasing the intensity of the cross-correlated light LT CC  generated by the nonlinear optical effect in the member  6 . This is based on the fact that the intensity of the cross-correlated light LT CC  varies proportional to the intensity of the incoming light LT O  and LT S  if the member  6  is made of material representing the secondary nonlinear optical effect. Also, this is based on the fact that the intensity of the cross-correlated light LT CC  varies proportional to the square of the intensity of the incoming light LT O  and LT S  if the member  6  is made of material representing the tertiary nonlinear optical effect.  
         [0218]    As a result, the apparatus  100 D according to the seventh embodiment of FIG. 10 has an additional advantage that the measurement sensitivity is raised compared with the apparatus  100  according to the first embodiment.  
         [0219]    For example, when the wavelengths of the target light LT O  and the sampling light LT S  is in the 1.3 μm band and the 1.5 μm band, respectively, an optical fiber amplifier doped with a rare-earth element or a semiconductor optical amplifier may be used as the optical amplifiers  54   a  and  54   b . In this case, he amplifiers  54   a  and  54   b  emit undesired, amplified optical components (e.g., light due to spontaneous emission) and thus, it is preferred that an optical band-pass filter is additionally provided to selectively pass the target light LT O  and the sampling light LT S . If so, the measurement sensitivity is raised furthermore.  
       EIGHTH EMBODIMENT  
       [0220]    [0220]FIG. 11 shows an apparatus  100 E for measuring the waveform of sampling light according to an eighth embodiment of the invention, which has the same configuration as the apparatus  100  according to the first embodiment except that an optical amplifier  55  is additionally provided. The optical amplifier  55  is located between the nonlinear optical member  6  and the optical filter  7 . Therefore, the explanation about the same configuration is omitted here for the simplification by attaching the same reference symbols as used in the first embodiment in FIG. 11.  
         [0221]    With the apparatus  100 E according to the eighth embodiment of FIG. 11, the amplified cross-correlated light LT CC  and the amplified background light LT BA  outputted by the optical amplifier  55  enters the optical filter  7 . Thus, the amplified cross-correlated light LT CC  is applied to the photodetector  8 , which means that the intensity of the light LT CC  is enhanced. As a result, the apparatus  100 E according to the eighth embodiment has an additional advantage that the measurement sensitivity is raised compared with the apparatus  100  according to the first embodiment of FIG. 2.  
       NINTH EMBODIMENT  
       [0222]    [0222]FIG. 12 shows an apparatus  100 F for measuring the waveform of sampling light according to a ninth embodiment of the invention, which has the same configuration as the apparatus  100  according to the first embodiment except that an electrical amplifier  56  is additionally provided. The electrical amplifier  56  is located between the photodetector  8  and the signal processing circuit  9 . Therefore, the explanation about the same configuration is omitted here for the simplification by attaching the same reference symbols as used in the first embodiment in FIG. 12.  
         [0223]    With the apparatus  100 F according to the ninth embodiment of FIG. 12, the electrical pulsed signal S CC  outputted by the photodetector  8  is amplified by the amplifier  56 , thereby supplying the amplified electrical signal S CCA  to the signal processing circuit  9 . As a result, the apparatus  100 F according to the eighth embodiment has an additional advantage that the measurement sensitivity is raised compared with the apparatus  100  according to the first embodiment.  
       TENTH EMBODIMENT  
       [0224]    [0224]FIG. 13 shows an apparatus  100 G for measuring the waveform of sampling light according to a tenth embodiment of the invention, which has the same configuration as the apparatus  100  according to the first embodiment except that a frequency-divider  57  is additionally provided. The frequency-divider  57  is located between the PLL circuit  4  and the signal processing circuit  9 . Therefore, the explanation about the same configuration is omitted here for the simplification by attaching the same reference symbols as used in the first embodiment in FIG. 13.  
         [0225]    With the apparatus  100  according to the first embodiment of FIG. 2, the electrical driving signal S D  from the PLL circuit  4  is directly supplied to the signal processing circuit  9  and therefore, the signal S D  itself is used for sampling in the circuit  9 . Unlike this, with the apparatus  100 G according to the tenth embodiment of FIG. 13, the driving signal S D  is frequency-divided by the frequency-divider  57  and therefore, the frequency-divided driving signal S DD  is used for sampling in the circuit  9 .  
         [0226]    For example, if the frequency of the driving signal S D  is divided by the factor of  10 , the count of the peak values of the cross-correlated signal LT CC  represented by the circular dots in FIG. 14 is decreased to one-tenth (i.e., {fraction (1/10)}). This means that the sampling rate is lowered to one-tenth compared with the first embodiment.  
         [0227]    The lowering of the sampling rate in the signal processing circuit  9  produces an additional advantage that the load applicable to the conversion operation from analog signals to digital ones in the circuit  9  can be reduced compared with the apparatus  100  according to the first embodiment.  
       VARIATIONS  
       [0228]    In the above-describe first to tenth embodiments of the invention, a passive mode-locked semiconductor laser  13  is used to configure the optical clock extractor  1 . However, the invention is not limited to this. Any other mode-locked laser may be used for this purpose.  
         [0229]    In the above-described embodiments, the invention is applied to measurement of the waveform of ultra-high speed optical pulses having a data transmission rate that exceeds about 40 Gb/s in each wavelength in real time with sufficiently high time resolution. However, the invention is not limited to this case. Needless to say, the invention may be applied to cases where the measuring apparatus is electrically connected to the external apparatus by way of a comparatively short optical transmission path, or to cases where the waveform of optical pulses having a data transmission rate that equal to or less than about 40 Gb/s in each wavelength is measured in real time with sufficiently high time resolution.  
         [0230]    Moreover, the PLL circuit  31  or  41  used in the second or third embodiment may be used in any one of the fourth to tenth embodiments. If the elements or devices additionally provided in the fourth to tenth embodiments are suitably combined together, any other embodiments maybe created easily. Needless to say, these embodiments thus created are included in the invention.  
         [0231]    While the preferred forms of the present invention have been described, it is to be understood that modifications will be apparent to those skilled in the art without departing from the spirit of the invention. The scope of the present invention, therefore, is to be determined solely by the following claims.