Abstract:
A fault current tolerant power supply circuit having a dc power source and switching devices for inverting dc current from the source into ac load current includes a protective circuit for protecting the switching devices from overvoltage conditions when an inductive load is short-circuited. The protective circuit uses a capacitive component to absorb charge during a short circuit to protect non-conducting switching devices from an overvoltage condition.

Description:
CROSS REFERENCE TO RELATED APPLICATIONS 
     This application claims the benefit of U.S. Provisional Application No. 60/311,822, filed Aug. 13, 2001. 
    
    
     FIELD OF THE INVENTION 
     The present invention relates to a power supply circuit that is of particular use for providing ac current to inductive loads as used, for example, in induction heating or melting applications wherein the inductive load may be subject to a short circuit that can produce, depending upon the topology of the supply, a high level of fault voltage across, or fault current through, switching devices used in the supply. 
     BACKGROUND OF THE INVENTION 
     A conventional series-resonant dc-to-ac inverter power supply circuit for an induction furnace, or other load having a substantially inductive impedance, includes a dc power source and an inverter having one or more pairs of series-connected switch circuits. Each switch circuit comprises a series combination of an inductive reactor and switching device, such as a silicon-controlled rectifier (SCR), with an antiparallel diode connected across the switching device. The inductive reactor limits the rate of change of current over time through its associated SCR when it turns-on (commutates) and, consequently, is referred to as a di/dt reactor. The inverter&#39;s SCRs are connected to an inductive load, such as an induction coil, and are alternatively gated (triggered) to switch between the non-conducting and conducting states. With this arrangement, each SCR in a pair of SCRs alternately allows current to flow through the induction coil in opposing directions, thus establishing an ac current flow through the coil. 
     Parallel-resonant dc-to-ac inverter power supply circuits are also used with inductive loads. However, series-resonant dc-to-ac inverter power supplies are preferred because of their superior controllability. Series-resonant dc-to-ac inverter power supplies are susceptible to short circuits in the inductive load. For example, when the load is an induction coil used in an electric induction furnace for metal melting, it is not unusual for spilled molten metal, or scrap metal being loaded into the furnace, to come into contact with the coil and cause at least a partial short circuit between two or more of the coil turns. The resulting instantaneous over-voltage condition across a non-conducting switching device in the inverter at the time of the short circuit can degrade or destroy the device. A known solution to the problem is to trigger the non-conducting switching device into conduction to eliminate the over-voltage condition across the device. However, a disadvantage of this approach is that it causes extremely high current to flow through the switching device which, in turn, generates significant heat in the device over a very short period of time. The switching devices are, in effect, forced to withstand the extremely high current in order to avoid being subjected to the over-voltage condition. The result of subjecting the switching device to these high current levels is degradation of the device and premature failure. 
     A solution to this problem is disclosed in U.S. Pat. No. 6,038,157. This patent teaches over-voltage protection of switching devices by adding a protective inductor in series with the load induction coil as illustrated in attached FIG.  1 . Circuit  110  includes a dc power source comprising a rectifier bridge circuit  120  (shown diagrammatically), series filter inductor  174  and parallel filter capacitor  172 , and two solid state switching device  130 , arranged in inverse parallel configuration. Each switching device has one terminal connected to an output bus of the de power source. Antiparallel diode  132  is connected across each switching device. A suitable, but non-limiting, switching device is a gate-controlled semiconductor device, such as an SCR. A di/dt reactor  140  is connected in series between the pair of switching devices as shown in FIG.  1 . Protective coil  150  has a first terminal connected to the approximate electrical midpoint of di/dt reactor  140 , and a second terminal connected to a first terminal of load induction coil  160 . The second terminal of load induction coil  160  is connected to the common connection between two series-connected commutation or tank capacitors  170  which, in series combination, are connected across the output buses of the dc power source. In an induction metal melting application, load induction coil  160  is typically wound around the exterior of the heating crucible. Direct current supplied from the power source is positively and negatively switched through switching devices  130  to supply an ac current to load induction coil  160 . The current flowing through coil  160  generates a magnetic field that inductively couples with a metal load in the crucible. The magnetic field induces an eddy current in the metal load that heats the metal. Since protective coil  150  continuously carries full load current, it generates significant power losses that decrease the overall efficiency of the power supply circuit. Furthermore the losses increase with the switching frequency of the supply. In the event of a short circuit in load induction coil  160 , the voltage applied to switching devices  130  is reduced by a voltage dividing circuit that comprises protective coil  150  and di/dt reactor  140 . 
     The present invention solves the problem of premature failures of the switching devices from exposure to over-voltage conditions resulting from short circuits in the load induction coil without penalizing circuit efficiency under normal operation, and without subjecting the switching devices to high level of currents to avoid the over-voltage. 
     BRIEF SUMMARY OF THE INVENTION 
     In one aspect, the present invention is a fault tolerant power supply circuit for an inductive load that protects sensitive power switching devices from excessive over-voltage conditions by straddling a pair of switching devices in the leg of an inverter circuit with a protective capacitive element. The protective capacitive element suppresses an over-voltage that would otherwise be applied across the switching devices in the event that a short circuit occurs in the load circuit. 
     In another aspect, the present invention is a fault tolerant power supply circuit that comprises a protective circuit for preventing the voltage across a non-conducting switching device from exceeding the peak allowable voltage of the non-conducting switching device during a short circuit in an inductive load. The protective circuit comprises a series connected blocking diode and protective capacitor, and a discharge resistor. The series combination of the blocking diode and protective capacitor is connected across the series combination of a pair of switch circuits. Each switch circuit comprises a switching device connected anti-parallel to an antiparallel diode. The discharge resistor may be connected across the protective capacitor or between the common connection of the series connected blocking diode and the protective capacitor, and the positive dc bus. A protective circuit can be used for each pair of switch circuits in power supplies with multiple pairs of switch circuits, such as full-bridge inverters. 
     In another aspect, the present invention is a fault tolerant power supply circuit that comprises a protective circuit for preventing the voltage across a non-conducting switching device from exceeding the peak allowable voltage of the non-conducting switching device during a short circuit in an inductive load. The protective circuit also clamps voltage overshoots across a switching device when an antiparallel diode transitions to reverse bias. The protective circuit comprises a series connected blocking diode and protective capacitor, and a series connected discharge resistor and choke. The discharge resistor is connected between the common connection of the series connected blocking diode and the protective capacitor, and the positive dc bus via the choke. A protective circuit can be used for each pair of switch circuits in power supplies with multiple pairs of switch circuits, such as full-bridge inverters. 
     Other aspects of the invention are set forth in this specification and the appended claims. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For the purpose of illustrating the invention, there is shown in the drawings a form that is presently preferred; it being understood, however, that this invention is not limited to the precise arrangements and instrumentalities shown. 
     FIG. 1 is a simplified schematic diagram of a prior art fault tolerant power supply circuit. 
     FIG.  2 ( a ) is a simplified schematic diagram of one example of the fault tolerant power supply circuit of the present invention wherein a half-bridge inverter is used. 
     FIG.  2 ( b ) is a simplified schematic diagram of another example of the fault tolerant power supply circuit of the present invention wherein a full-bridge inverter is used. 
     FIG.  2 ( c ) is a simplified schematic diagram of another example of the fault tolerant power supply circuit of the present invention wherein a half-bridge inverter is used. 
     FIG.  2 ( d ) is a simplified schematic diagram of another example of the fault tolerant power supply circuit of the present invention wherein a full-bridge inverter is used. 
     FIG. 3 is a simplified equivalent schematic diagram of one example of the fault tolerant power supply circuit of the present invention immediately following the occurrence of a short circuit across an induction coil in the circuit when SCR  30   b  in FIG.  2 ( a ) is non-conducting. 
     FIG.  4 ( a ) is a waveform diagram illustrating the short circuit protection characteristics achieved with the example of the fault tolerant power supply circuit of the present invention shown in FIG.  2 ( a ) or FIG.  2 ( c ). 
     FIG.  4 ( b ) is a waveform diagram illustrating the short-circuit characteristics of a power supply circuit without the protection provided by the fault tolerant power supply circuit of the present invention shown in FIG.  2 ( a ) or FIG.  2 ( c ). 
     FIG.  5 ( a ) is a simplified schematic diagram of another example of the fault tolerant power supply circuit of the present invention wherein a half-bridge inverter is used. 
     FIG.  5 ( b ) is a simplified schematic diagram of another example of the fault tolerant power supply circuit of the present invention wherein a full-bridge inverter is used. 
     FIG. 6 is a waveform diagram illustrating the voltage clamping characteristics achieved with the example of the fault tolerant power supply circuit of the present invention shown in FIG.  5 ( a ). 
     FIG. 7 is a graphical illustration of the method of selecting the capacitance of a protective capacitor used in the fault tolerant power supply circuit of the present invention. 
    
    
     DETAILED DESCRIPTION OF THE INVENTION 
     Referring to the drawings, wherein like numerals indicate like elements, there is shown in FIG.  2 ( a ) one example of the fault tolerant power supply circuit  10  of the present invention. Each of two switching devices  30   a  and  30   b  (such as, but not limited to SCRs) is connected anti-parallel with antiparallel diodes  32   a  and  32   b , respectively, to form switch circuits  31   a  and  32   b  respectively. The first terminals of the switch circuits are connected together at switch common connection  93 . The first terminals of di/dt reactors  40   a  and  40   b  are connected to the second terminals of switch circuits  31   a  and  31   b , respectively, to form switch modules  37   a  and  37   b , respectively. The second terminals of di/dt reactors  40   a  and  40   b  are connected to the positive and negative dc buses (output lines) of dc power source  20 , respectively. The power source comprises rectifier bridge  20  (shown diagrammatically) with optional series filter inductor  74  and parallel filter capacitor  72 . 
     Protective circuit  55  comprises a series-connected blocking diode  52  and protective capacitor  54  with common connection  95 , and discharge resistor  53  connected in parallel with capacitor  54 , as shown in FIG.  2 ( a ). The capacitance of protective capacitor  54  is selected as further described below so that it will charge to a peak voltage value that is less than the peak allowable voltage for the switching device. The peak allowable voltage for the switching device may be the rated repetitive forward blocking voltage of the device, which is the maximum instantaneous value of forward voltage that occurs across the device. Lines  90  and  91  of protective circuit  55  are connected between the first terminals of di/dt reactors  40   a  and  40   b  (second terminals of switch circuits  31   a  and  31   b ), respectively. 
     First terminal of load induction coil  60  is connected to switch common connection  93 . The second terminal of load induction coil  60  is connected to a capacitor common connection  94  between two series-connected commutation or tank capacitors  70   a  and  70   b , that, in series combination, are connected across the positive and negative dc buses, respectively, of the power source. The commutation capacitors divert load current from a switching device until the current drops below the holding current and causes the switching device to turn off. Relative to the load induction coil, the pair of switching modules are connected in inverse parallel configuration. 
     When di/dt reactors are used, typically a snubber circuit is used across each switch circuit to control voltage transients across the switching device and antiparallel diode during commutation and voltage time rate-of-change (dv/dt) across the switching device. A typical, but non-limiting, example of an optional snubber circuit  65  is illustrated in FIG.  2 ( a ). The snubber circuit comprises capacitor  36 , resistor  34  and diode  33  connected together in series, and a resistor  35 , connected in parallel across the series combination of diode  33  and resistor  34 . Capacitor  36  and resistor  34  form a series resonant circuit with the di/dt reactor in its associated switch module. Resistor  35  generally has a resistance value much greater than that of resistor  34  to decrease the discharge current of capacitor  36  through the switching device in the associated switch circuit. 
     In an induction metal melting application, load induction coil  60  is typically wound around the exterior of the crucible in which the metal load is placed. By alternate cycling of switching devices  30   a  and  30   b , dc output current from power source  20  establishes an ac current through load induction coil  60 . The ac current flowing through the coil generates a magnetic field that inductively couples with the metal load in the crucible. The magnetic field generates an eddy current in the metal load which heats and melts the metal. 
     When the power supply circuit  10  is in operation, protective capacitor  54  charges up to a peak voltage that is somewhat higher than the de bus voltage and less than the peak allowable voltage of switching devices  30   a  and  30   b . Dc charge current is provided through diode  52  from power source  20 , and during the transitions of the antiparallel diodes from forward to reverse bias. Impedance values for resistor  53  and capacitor  54  are selected so that the R-C time constant for the circuit is sufficiently large in comparison with the time period of the operating frequency as established by the switching frequency of devices  30   a  and  30   b . Generally, to meet this condition and minimize power losses, resistor  53  should be sufficiently large to provide a complete discharge time of at least 60 seconds for capacitor  54 . This time should not exceed the time for discharging filter capacitor  72 , if used. Additionally, the inverse of the product of the resistance and capacitance of resistor  53  and capacitor  54 , respectively, will be much greater than the operating switching frequency of the switch circuits. 
     As an example of the operation of protective circuit  55 , a worst-case condition of an unintentional shorting of load induction coil  60  occurs shortly after the current flow through switching device  30   a  has crossed zero reference and antiparallel diode  32   a  has started to conduct. At this time, the voltages across load induction coil  60  and commutation capacitor  70   b  are at their maximum values. The equivalent circuit for these conditions is shown in FIG. 3, with coil  60  shorted (not shown) and switching device  30   b  non-conducting. 
     The maximum voltage across commutation capacitor  70   b  is applied across diodes  32   a  and  52 , protective capacitor  54  and di/dt reactor  40   b . Since the voltage across a capacitor cannot change instantaneously, the voltage difference between capacitor  70   b  and protective capacitor  54  must be instantaneously applied across di/dt reactor  40   b . Non-conducting switching device  30   b , connected in parallel with protective capacitor  54  (via diodes  32   a  and  52  with negligible voltage drop when forward biased and conducting current) will be subject to the same voltage as that across protective capacitor  54 . The capacitance of protective capacitor  54  is selected so that the short circuit current can not charge capacitor  54  to a voltage that exceeds the peak allowable voltage for switching devices  30   a  and  30   b . Consequently, the capacitance of protective capacitor  54  will be sufficient to absorb a charge from commutation capacitors  70   a  and  70   b  to a voltage that is less than the peak allowable voltage rating of switching devices  30   a  and  30   b.    
     FIG.  2 ( b ) illustrates a similar protection scheme as that in FIG.  2 ( a ) for an inverter with a full-bridge rectifier rather than a half-bridge rectifier. A single commutation capacitor  70  can be used in series with load induction coil  60  across the ac output of the full-bridge. A protective circuit  55  is provided for each of the two pairs of switch modules that comprise the two legs of the full-bridge inverter. 
     FIG.  2 ( c ) illustrates another example of the fault tolerant power supply circuit of the present invention wherein the first terminal of discharge resistor  53  in protective circuit  55   a  is connected to common connection  95 , and the second terminal of discharge resistor  53  is connected to the positive dc bus of power source  20  so that protective capacitor  54  discharges through the positive dc bus. FIG.  2 ( d ) illustrates a similar protection scheme as that in FIG.  2 ( c ) for an inverter with a full-bridge rectifier rather than a half-bridge rectifier. A single commutation capacitor  70  can be used in series with load induction coil  60  across the ac output of the full-bridge. A protective circuit  55   a  is provided for each of the two pairs of switch modules that comprise the two legs of the full-bridge inverter. 
     FIG. 7 illustrates one method for determining suitable values of capacitance (C 54 ) for protective capacitor  54 . The y-axis in FIG. 7 represents the peak allowable voltage (V pk ) across switching devices  30   a  and  30   b  defined as the ratio of the peak allowable voltage to the dc voltage (E) of the de power source. The x-axis in FIG. 7 represents the appropriate selection of capacitance C 54  as the ratio of capacitance C 54  to the equivalent commutation or tank capacitance (C com ), which is equal to the sum of the capacitances for commutation capacitors  70   a  and  70   b , or the capacitance of the single commutation capacitor  70  for a full-bridge inverter. Curves  81 ,  82  and  83  in FIG. 7 are representative of a family of curves wherein each curve is defined by a unique value of the ratio (R) of the voltage (V tcap ) across either capacitor  70 ( a ) or capacitor  70 ( b ) (substantially equal voltages), or capacitor  70  for a full-bridge circuit, to the voltage E of the de power source (R=V tcap /E). Further ratio R is increasing in value for curves in the direction from curve  81  to curve  83 . Illustrating the method of using FIG. 7 to select a suitable value for capacitance C 54 , assume “S” represents the allowed y-axis ratio for peak allowable voltage (V pk ), for example 1,800 volts. Then the x-axis ratio for C 54  must be selected from the portion of the appropriate curve (determined by the fixed operating voltage V tcap  for a particular application) in the shaded area below the dash line “S”. Dc power source voltage E is substantially constant for a particular operating system. 
     Major factors impacting the desired capacitance of protective capacitor  54  are: the maximum peak allowable voltage of switching devices  30   a  and  30   b ; peak voltage observed across the commutation capacitors; voltage across protective capacitor  54  immediately prior to a short circuit in load induction coil  60 ; and the capacitance of the commutation capacitors. For example, if: the voltage across protective capacitor  54  immediately prior to a short circuit in coil  60  is 1,000 volts; the maximum peak allowable voltage for switching devices  30   a  and  30   b  is 1,800 volts; and the peak voltage observed across the commutation capacitors is 4,200 volts, the desired capacitance of protective capacitor  54  is at least five times the equivalent commutation capacitance. 
     FIG.  4 ( a ) and FIG.  4 ( b ) graphically illustrate the advantage of the protective circuit  55  of the present invention as shown in FIG.  2 ( a ) through FIG.  2 ( d ). Curve V 70b  represents the voltage over time (in milliseconds (ms) for this particular example) across commutation capacitor  70   b . Curve V 60  represents the voltage over time across load induction coil  60 . Curve V 30b  represents the voltage over time across switching device  30   b . Curve V 30a  represents the voltage over time across switching device  30   a . Curve V 54  represents the voltage over time across protective capacitor  54 . An unintentional short-circuit occurs at the time marked as “SC” in FIG.  4 ( a ) and FIG.  4 ( b ). In FIG.  4 ( b ) (no protective capacitor  54 ), at time SC, the voltage across switching device  30   b  rises to almost 4,000 volts. In FIG.  4 ( a ) (with protective capacitor  54 ), at time SC, the voltage across switching device  30   b  is held to approximately 1,450 volts, which is the voltage on protective capacitor  54 , and is far less than the 1,800 volts maximum allowable peak voltage for switching devices  30   a  and  30   b . Prior to the time of the short circuit, protective capacitor  54  is charged to approximately 1,000 volts. The voltage on protective capacitor  54  rises an additional 450 volts due to charge from commutation capacitor  70   b.    
     FIG.  5 ( a ) illustrates another example of a fault tolerant power supply circuit of the present invention. In this example, snubber circuits  65   a  do not use charging resistor  34  as in the examples shown in FIG.  2 ( a ) through FIG.  2 ( d ). Protective circuit  55   b  in FIG.  5 ( a ) includes a series-connected diode  52   a  and protective capacitor  54   a  with common connection  95   a . The first terminal of discharge resistor  53   a  is connected to common connection  95   a , and the second terminal of resistor  53   a  is connected to the first terminal of choke  56 , which has its second terminal connected to the positive dc bus of the dc power source. Lines  90   a  and  91   a  of protective circuit  55   b  are connected between the first terminals of di/dt reactors  40   a  and  40   b  (second terminals of switch circuits  31   a  and  31   b ), respectively. 
     Snubber charging resistor  34  shown in FIG.  2 ( a ) through FIG.  2 ( d ) is used to dampen the oscillation in the series-resonant circuit formed by the associated di/dt reactor and snubber capacitor  36 . This reduces the voltage spike on the associated switching device at the time when the antiparallel diode in the other switch circuit transitions to reverse bias. However use of charging resistor  34  has negative effects since it increases the non-conducting dv/dt of its associated switching device and generates power losses that increase as the switching frequency increases. 
     The fault tolerant protection power supply circuit shown in FIG.  5 ( a ) can be used without a snubber charging resistor since protective circuit  55   b  is connected across switching devices  30   a  and  30   b . Protective capacitor  54   a , if selected as disclosed above, will have sufficient capacitance to clamp voltage overshoots across a switching device at the time when the antiparallel diode in the other switch circuit transitions to reverse bias. FIG. 6 graphically illustrates the advantage of protective circuit  55   b  as shown in FIG.  5 ( a ). Curve V 54a  represents the voltage over time (in microseconds (μs) for this particular example) across charged protective capacitor  54   a , which in this example is 1,000 volts. Curve V′ 30  represents the voltage over time across switching device  30   a  or  30   b  for the circuits shown in FIG.  2 ( a ) through FIG.  2 ( d ). Curve V 30  represents the voltage over time across switching device  30   a  or  30   b  for the circuit shown in FIG.  5 ( a ). Curve  154   a  represents the current over time through protective capacitor  54   a  in FIG.  5 ( a ). As illustrated by curve V 30 , the voltage across switching device  30   a  or  30   b  in FIG. 5 will be clamped to the voltage of protective capacitor  54   a . Comparing this with curve V′ 30 , which represents the voltage across a switching device in a circuit utilizing a snubber charging resistor as shown in FIG.  2 ( a ) through FIG.  2 ( d ), the dv/dt is greater for the circuit with the snubber charging resistor than without it since the difference in voltage curves V 30  and V′ 30  is equal to the voltage drop on charging resistor  34 . 
     Since the pair of antiparallel diodes used in the circuit of FIG.  5 ( a ) switch from forward bias (conduction) to reverse bias (non-conduction) at the switching frequency for the pair of switching devices in the circuit, clamping V 30  will result in further charging of protective capacitor  54   a . This additional charge or voltage on protective capacitor  54   a  will be discharged via resistor  53   a  and choke  56 . The resistance of resistor  53   a  and inductance of choke  56   a  are selected to optimize the return of energy back to the dc power source from snubber circuit  65   a  and maintain the appropriate clamp voltage. Further the use of resistor  53   a  in series with choke  56  will reduce the Root Mean Square (RMS) value and jitter of the discharge current, I 54a , through protective capacitor  54   a . Consequently, power loss in discharge resistor  53   a  is reduced. If choke  56  is fabricated to have a suitable value of resistance, resistor  53  and choke  56  may be replaced by a single resistive/inductive element. 
     If snubber circuit  65   a  shown in FIG.  5 ( a ) has components selected to maintain the same dv/dt value for snubber circuit  65  in FIG.  2 ( a ) through FIG.  2 ( d ), the capacitance of snubber capacitor  36  in FIG.  5 ( a ) will be less than the capacitance of snubber capacitor  36  in FIG.  2 ( a ) through FIG.  2 ( d ). Consequently, the power loss in snubber resistor  35  in FIG.  5 ( a ) will be less than that for snubber resistor  35  in FIG.  2 ( a ) through FIG.  2 ( d ). 
     FIG.  5 ( b ) illustrates a similar protection scheme as that in FIG.  5 ( a ) for an inverter with a full-bridge rectifier rather than a half-bridge rectifier. A single commutation capacitor  70  can be used in series with load induction coil  60  across the ac output of the full-bridge. A protective circuit  55  is provided for each of the two pairs of switch modules that comprise the two legs of the full-bridge rectifier. 
     The examples of the invention include reference to specific electrical components. One skilled in the art may practice the invention by substituting components that are not necessarily of the same type but will create the desired conditions or accomplish the desired results of the invention. For example, single components may be substituted for multiple components or vice versa. Further one skilled in the art may practice the invention by rearranging components to create the desired conditions or accomplish the desired results of the invention. While a single pair of switch circuits are illustrated in the examples of the invention, multiple pairs of switch circuits connected in parallel may be utilized to practice the present invention. The short circuit on the load induction coil may be a partial short circuit of the coil in all of the examples of the invention. Further while the examples illustrate operation of the invention in series-resonant power supplies, the invention is applicable to other power supply topologies with appropriate modifications as understood by one who is skilled in the art. 
     The foregoing examples do not limit the scope of the disclosed invention. The scope of the disclosed invention is further set forth in the appended claims.