Abstract:
A method for modulating the E-field of an optical carrier signal utilizes a Mach-Zehnder modulator having a pair of independently controllable branches. A pair of independent branch drive signals V L (t) and V R (t) are derived. Each branch of the MZ modulator is driven with a respective one of the independent branch drive signals. By this means, a low cost conventional MZ modulator (interferometer) can be used to perform complex modulation of the E-field of the optical carrier. In some embodiments, this functionality is used to facilitate precompensation of optical impairments of an optical communications system.

Description:
CROSS-REFERENCE TO RELATED APPLICATIONS 
     This application is a Continuation-in-Part of Applicant&#39;s co-pending U.S. patent applications Ser. Nos. 10/307,466 filed Dec. 2, 2002; and Ser. No. 10/405,236 filed Apr. 3, 2003. This application also claims priority of International Patent Application No. PCT/CA03/01044 filed Jul. 11, 2003. 
    
    
     MICROFICHE APPENDIX 
     Not Applicable. 
     TECHNICAL FIELD 
     The present invention relates to optical signal transmitters for optical communications systems, and in particular to optical E-field modulation using a Mach-Zehnder (MZ) interferometer. 
     BACKGROUND OF THE INVENTION 
     In the optical communications space, various techniques are used to synthesize an optical communications signal for transmission. A popular technique utilizes a laser  2  coupled to an external optical modulator  4 , as shown in  FIG. 1   a . The laser  2  generates a narrow-band continuous wave (CW) optical carrier signal  6  having a desired wavelength. The optical modulator  4  operates to modulate the amplitude and/or phase the carrier signal  6  to generate the optical communications signal  8  based on a drive signal  10  that encodes data to be transmitted. Typically, the drive signal  10  is generated by a driver circuit  12  based on an input data signal x(t). 
     In the arrangement illustrated in the  FIGS. 1   a - 1   b , the optical modulator  4  is provided by a well known Mach-Zehnder (MZ) interferometer. Other types of modulators may be used, depending on the desired type of modulation. For example, an electro-absorptive modulator (EAM) or a variable optical attenuator (VOA) may be used for amplitude modulation; whereas phase shifters are well known for implementing phase modulation. In each case, the driver circuit  12  generates the drive signal  10  by scaling the input data signal x(t) to satisfy the voltage and current requirements of the modulator  4 . The driver circuit  12  may also generate one or more bias signals (not shown) for controlling a bias point of the modulator  4  in a manner well known in the art. 
     In the embodiment of  FIG. 1   a , the MZ modulator is controlled by a single drive signal  10  V(t), which, in this case, would typically be a scaled version of the input data signal x(t). In  FIG. 1   b , a “dual-branch” MZ modulator  4  is controlled by a pair of differential drive signals +V(t) and −V(t), which are substantially equal and opposite. The use of a differential pair of drive signals has the well known advantage that a desired amplitude modulation of the communications signal  8  can be obtained with drive signal voltage levels that are approximately half of that required for the single-drive embodiment of  FIG. 1   a . Additionally, it is possible to minimize undesirable signal chirp by adjusting the scaling factors applied to each of the differential drive signals +V(t) and −V(t) 
     In either of the above embodiments, the MZ modulator displays a sinusoidal response to the applied drive voltage (or voltage difference in the case of differential drive signals) as shown in  FIG. 1   c . Thus, a DC bias signal (not shown) is used to drive the modulator  4  to a desired bias point  14  of the response curve  16 . The drive signal(s)  10  can then drive excursions of the modulator response, corresponding to maximum and minimum transmittance, respectively. This operation yields an amplitude-modulated optical communications signal  8  which carries the original data signal x(t), and is typically employed to implement On-Off-Keying (OOK) transmission protocols. 
     As is well known in the art, a directly analogous approach can be employed using a phase shifter as the optical modulator  4 . In this case, the drive signal(s)  10  drive excursions of the modulator phase response, which yields a phase-modulated communications signal  8 . This technique is typically used to implement Phase Shift Keying (PSK) or, more commonly Differential Phase Shift Keying (DPSK) transmission protocols. 
     A limitation of the optical signal synthesizers illustrated in  FIGS. 1   a – 1   c , is that they are designed to modulate only a single dimension (i.e. amplitude or phase) of the CW optical carrier signal  6  generated by the narrow-band laser  2 . However, in some instances it is desirable to modulate two or more dimensions of the CW carrier  6 . For example, Applicant&#39;s co-pending U.S. patent application Ser. Nos. 10/262,944, filed Oct. 3, 2002; Ser. No. 10/307,466 filed Dec. 2, 2002; and Ser. No. 10/405,236 filed Apr. 3, 2003; and International Patent Application No. PCT/CA03/01044 filed Jul. 11, 2003 describe techniques for compensating impairments in an optical link by predistoring an input signal, in the electrical domain, and then using the thus predistorted signal to drive the optical modulator. As described in those applications, successful implementation of this technique, particularly for the case of polarisation dependent and non-linear impairments, requires the use of an optical modulator capable of modulating both the amplitude and phase of the CW carrier  6 . Stated more generally, it is desirable to modulate the E-field of the CW carrier, within the complex plane. 
     Various methods of accomplishing this result are described in Applicant&#39;s co-pending U.S. patent application Ser. Nos. 10/262,944, filed Oct. 3, 2002; Ser. No. 10/307,466 filed Dec. 2, 2002; and Ser. No. 10/405,236 filed Apr. 3, 2003; and International Patent Application No. PCT/CA03/01044 filed Jul. 11, 2003. As shown in  FIGS. 2   a  and  2   b , most of these techniques utilize multiple one-dimensional modulators in combination. Thus, for example,  FIG. 2   a  shows a complex Mach-Zehnder modulator  18  composed of two 1-D MZ modulators  4   a ,  4   b  connected in parallel. A complex driver circuit  20  generates the drive signal(s)  10  in the form of In-phase and Quadrature signal components V I (t), V Q (t), each of which may be represented by a single signal or a differential signal pair. The drive signals  10  are used to drive a respective one of the branch MZ modulators  4   a ,  4   b . This technique enables arbitrary E-field modulation of the CW carrier  6  throughout the complex I-Q plane. 
     In  FIG. 2   b , a conventional 1-D MZ modulator  4  is cascaded with a phase shifter  22 . In this case, the complex driver circuit  20  generates the drive signals  10  in the form of amplitude and phase signal components V S (t) and V φ (t) (each of which may be represented by a single signal or a differential signal pair), which are respectively used to drive the MZ modulator  4  and the phase shifter  22 . This technique enables arbitrary E-field modulation of the CW carrier throughout the complex polar-coordinate (Amplitude-Phase) plane. 
     In  FIG. 2   c , the frequency dependence of conventional lasers on the drive current is used in conjunction with a conventional 1-D MZ modulator  4 . In this case, the complex driver circuit  20  generates the drive signal(s)  10  in the form of amplitude and frequency signal components V S (t) and V f (t). The amplitude component V S (t) (which may be represented by a single signal or a differential signal pair) drives the MZ modulator  4  to modulate the amplitude of the CW carrier signal  6  in a conventional manner. The frequency component V f (t) provides the laser drive current, and is varied to induce desired excursions of the laser frequency. This technique enables E-field modulation of the CW carrier within the complex polar-coordinate (Amplitude-Phase) plane, limited primarily by the frequency response of the laser  2 . 
     All of these prior solutions are advantageous in that they enable E-field modulation of the CW carrier  6 . However, the solutions of  FIGS. 2   a  and  2   b  are expensive, because multiple devices are required. The solution of  FIG. 2   c  requires only a single amplitude modulator  4 , but suffers a disadvantage that the laser is current-controlled. In some cases, obtaining the desired frequency modulation of the laser  2  may require very large and/or very rapid changes in the drive signal voltage. This increases the cost of the driver circuit  20 , and may result in the production of unwanted noise. 
     Accordingly, methods and apparatus for cost-effectively modulating the E-field of an optical carrier signal remains highly desirable. 
     SUMMARY OF THE INVENTION 
     Accordingly, an object of the present invention is to provide methods and apparatus for modulating the E-field of an optical carrier signal. 
     Thus, an aspect of the present invention provides a method for modulating the E-field of an optical carrier signal. According to the present invention, a Mach-Zehnder modulator having a pair of independently controllable branches is provided. A pair of independent branch drive signals V L (t) and V R (t) are derived. Each branch of the MZ modulator is driven with a respective one of the independent branch drive signals. By this means, a low cost conventional MZ modulator (interferometer) can be used to perform complex modulation of the E-field of the optical carrier. In some embodiments, this functionality is used to facilitate precompensation of optical impairments of an optical communications system. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Further features and advantages of the present invention will become apparent from the following detailed description, taken in combination with the appended drawings, in which: 
         FIGS. 1   a – 1   c  schematically illustrate principal components and operation of one-dimensional communications signal synthesizers known in the prior art; 
         FIGS. 2   a – 2   c  schematically illustrate principal components of complex communications signal synthesizers comprising multiple one-dimensional synthesizers of the type illustrated in  FIGS. 1   a – 1   c;    
         FIG. 3  schematically illustrates principal components of a complex communications signal synthesizer in accordance with an embodiment of the present invention; 
         FIG. 4  illustrates principal components and operation of the driver circuit usable in the embodiment of  FIG. 3 ; and 
         FIGS. 5   a – 5   d  schematically illustrate operation of the embodiment of  FIG. 3 . 
     
    
    
     It will be noted that throughout the appended drawings, like features are identified by like reference numerals. 
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     The present invention provides methods and apparatus for modulating the E-field of an optical carrier signal. Embodiments of the invention are described below, by way of example only, with reference to  FIGS. 3–5 . 
     For the purposes of understanding the present invention, it is useful to consider a representation of the total optical E-field E(t) as a vector confined to a plane and emanating from a fixed origin, where the length of the vector gives the amplitude of the E-field at any instant (t), and the direction of the vector gives the phase of the field at any instant (t). Within this construction, we consider two basis sets. The first basis set is a Cartesian coordinate system centered on the E-field origin. In this Cartesian representation, the total E-field E(t) is decomposed along the orthogonal h-Phase (I) and Quadrature (Q) direction. The second basis set is a polar coordinate system, again sharing its origin with that of the E-field vector. In this poles representation, the E-field is decomposed into vector length (S) and phase angle (φ) relative to the I direction, These two basis sets are related by a non-linear transformation, in a manner well known in the art. In each of these representations, the time-sequence of loci of the end-point of the E-field vector is referred to as a trajectory of the E-field 
     As shown in  FIG. 3 , a system in accordance with the present invention comprises a complex driver circuit  20  and a conventional “dual branch” MZ interferometer  4  of the type described above with reference to  FIG. 1   b . In operation, the complex driver circuit  20  generates a pair of independent drive signals V R (t) and V L (t), which are used to drive a respective branch  22  of the MZ interferometer  4 . This operation is designed to exploit the optical behaviour of the “dual branch” MZ interferometer, which is characterized by a respective independently controllable phase shifter  24  in each branch  22 . With this configuration, conventional amplitude modulation of the CW carrier  6  is obtained by driving each phase shifter  24  with respective equal and opposite drive signals  10 . This differential pair of drive signals produces corresponding equal and opposite phase delays experienced by light traversing the two branches  22   a, b  of the modulator  4 , via the well known electro-optical effect The optical power of the recombined lightwave (communications signal)  8  emerging from the MZ modulator  4  varies sinusoidally with the phase angle between the beams traversing the two branches  22 . 
     A trivial alternative mode of operation is to supply the same drive signal to both branches  22  of the “dual branch” MZ interferometer  4 . In this case, the differential phase delay will be zero, and the recombined signal a emerging from the modulator  4  will have a constant optical power. However, the (identical) voltage-dependent phase delay experienced by the light traversing the branches  22   a  and  22   b  of the modulator  4  will produce corresponding phase changes in the recombined signal  8 . 
     As may be appreciated, if the two branches  22  are independently driven (i.e. the two drive signals  10   a  and  10   b  are different) the recombined light wave  8  emerging from the MZ modulator  4  will exhibit excursions of both phase and amplitude. In the present invention, this behavior is exploited by deriving independent drive signals V R (t) and V L (t) which are calculated to yield desired excursions of the E-field of the recombined signal  8 . In very broad terms, this can be accomplished by representing the total E-field E(t) of the recombined signal  8  in polar coordinates: that is, in terms of orthogonal phase φ(t) and amplitude S(t) components. Based on the known spectral performance of the MZ modulator  4 , it is a simple matter to compute the voltage level V φ (t) which, when supplied to both branches  22  of the MZ modulator  4 , will produce the desired phase φ(t) component. Similarly, it is a simple matter to compute the differential voltage levels ±V S (t) which, when supplied to respective branches  22  of the MZ modulator  4 , will produce the desired E-field amplitude S(t) component. The required drive signals V R (t) and V L (t) can then be computed using equations of the form:
 
 V   L ( t )= V   φ ( t )+ V   s ( t )  (Eq. 1)
 
and
 
 V   R ( t )= V   φ ( t )− V   S ( t )  (Eq. 2)
 
     Those skilled in the art will appreciate that these computations can readily be accomplished using known digital signal processing techniques. Principal components and operation of a complex driver circuit  20  capable of performing this operation are described below with reference to  FIG. 4 . A practical application of the system of  FIG. 4 , for compensating dispersion, is also described with reference to  FIGS. 5   a – 5   d.    
     In the embodiment of  FIG. 4 , the complex modulator  20  comprises a digital filter  26  cascaded with a non-linear compensator  28 , which cooperate to generate successive digital sample values V R (n) and V L (n). These digital sample values are then converted into corresponding analog voltage levels by respective digital-to-analog converters (DACs)  30 , filtered (at  32 ) to reduce out-of-band noise, and scaled by low noise amplifiers  34  to yield the desired drive signals V R (t) and V L (t). If desired, respective digital filters  36  may be positioned between the non-linear compensator  28  and the DACs  30  in order to compensate any propagation delay differences between the DACs  30  and the MZ modulator  4 . 
     Various known digital filter types may be used to implement the digital filter  26 , such as, for example, Finite Impulse Response (FIR) filters, Infinite Impulse Response (IIR) filters, and Fast Fourier Transform (FFT filters). Alternatively, the digital filter  26  may be implemented using a Random Access Memory Look-up Table (RAM LUT). In either case, the digital filter  26  generates successive sample values I(n) and Q(n), which represent the desired E-Field in Cartesian (In-phase and Quadrature) coordinates, at a sample rate which is about double the bit-rate of the input signal x(t). In a trivial case, this operation can be provided as a simple mapping function, which maps the input signal x(t) ( FIG. 5   a ) onto the complex I-Q plane ( FIG. 5   b ) in which logical “0”s and “1”s of the input signal are shown mapped onto the I-axis. However, as described in applicant&#39;s co-pending U.S. patent application Ser. Nos. 10/262,944, filed Oct. 3, 2002; Ser. No. 10/307,466 filed Dec. 2, 2002; and Ser. No. 10/405,236 filed Apr. 3, 2003; and International Patent Application No. PCT/CA03/01044 filed Jul. 11, 2003, the digital filter  26  preferably also implements signal predistortion in accordance with a compensation function C(t) which substantially compensates impairments of an optical communications link (not shown). In this case, the E-field E(t) will exhibit arbitrary excursions in the I-Q plane, and sample values I(n) and Q(n) generated by the digital filter  26  may be markedly different from either previous or successive values, as may be seen in  FIG. 4   c  in which the shaded region represents the range of I(n) and Q(n) valves generated by the digital filter  26 . 
     The non-linear compensator  28  converts the Cartesian coordinate sample values I(n) and Q(n) into corresponding polar (amplitude and phase) coordinate values S(n) and φ(n). In order to limit the drive voltage requirements of downstream components (primarily the DACs  30 ), these polar coordinate values S(n) and φ(n) may be scaled and clipped to limit the phase φ(n) to within a selected range of angles, and to limit S(n) to an amplitude range corresponding to the minimum and maximum points of the modulator response curve  16 . For example, the phase φ(n) may be limited to a range of 90 degrees (i.e. ±45 degrees from the bias point) as shown in  FIG. 5   d . A wider or narrower angle range may be selected, based on the peak-to-peak voltage limitations of the driver circuit  20 . 
     In order to limit the bandwidth requirements of downstream components, complex E-field signal trajectories crossing very near the origin (which are equivalent to nearly instantaneous phase changes of 180 degrees) should be prevented by suitable scaling and clipping operations. For example, signal trajectories passing very near the origin can be eliminated by applying an offset to the signal such that it is moved away from the origin. In addition, a scaling factor may be applied to the Cartesian coordinate sample values I(n) and Q(n) so as to avoid maximum amplitude values of greater than 1 (that is, greater than the laser output). 
     In combination, the above-described scaling operations yield polar coordinate values S(n) and φ(n) concentrated within the dashed line of  FIG. 5   d . The shaded region of  FIG. 5   d  represents values that are attainable by the MZ modulator  4  when a drive signal peak-to-peak voltage limitation is in place, and the modulator is biased at an appropriate point of the response curve  16 . Polar coordinate values S(n) and φ(n) lying outside of the shaded region are removed by clipping, in a manner well known in the art. As may be appreciated, provided that the amount of such clipping is small, the resulting distortion of the E-field will not unduly disrupt operation of the system. 
     Once the polar coordinate values S(n) and φ(n) have been determined, the non-linear compensator  28  computes corresponding drive signal levels V S (n) and V φ (n) (e.g. using a simple look-up table function), and then calculates the required drive signal levels V L (n) and V R (n) using equations 1 and 2 above. 
     The embodiment(s) of the invention described above is(are) intended to be exemplary only. The scope of the invention is therefore intended to be limited solely by the scope of the appended claims.