Abstract:
An input circuit of a microwave amplification tube achieves improved instantaneous bandwidth. By directly coupling the transmission line carrying a modulating radio frequency signal to a control grid, a low-Q input circuit is created that increases the fractional bandwidth of the system. A resonant cavity may be used to generate a voltage across the gap between the cathode and the control grid. Alternative geometries are presented whereby the electron beam is emitted from a cathode connected either to the center conductor of the transmission line or to the outer conductor of the transmission line. Alternatively, the electric field of the radio-frequency signal propagating through the transmission line may be used to create a voltage across the gap between the cathode and the control grid without using a resonant cavity. Likewise, alternative geometries are presented by which the electron beam is emitted from a cathode connected either to the center conductor or to the outer conductor of the transmission line.

Description:
RELATED APPLICATION DATA 
       [0001]    This application claims the benefit, pursuant to 35 U.S.C. §119(e), of U.S. provisional application Ser. No. 60/867,756, filed Nov. 29, 2006. 
     
    
     BACKGROUND OF THE INVENTION 
       [0002]    1. Field of the Invention 
         [0003]    The present invention relates to microwave amplification tubes, such as an inductive output tube (IOT), and, more particularly, to an input circuit for an IOT or other emission-gated device providing improved instantaneous bandwidth. 
         [0004]    2. Description of Related Art 
         [0005]    It is well known in the art to utilize a linear beam device, such as a klystron or traveling wave tube amplifier, to generate or amplify a high-frequency RF signal. Such a device generally includes an electron-emitting cathode and an anode spaced therefrom. The anode includes a central aperture, and by applying a high voltage potential between the cathode and anode, electrons may be drawn from the cathode surface and directed into a high-power beam that passes through the anode aperture. One class of linear beam device, referred to as an inductive output tube (IOT), further includes a grid disposed in the inter-electrode region defined between the cathode and anode. The electron beam may thus be density modulated by applying a radio-frequency (RF) signal to the grid relative to the cathode. After the anode accelerates the density-modulated beam, the beam propagates across a gap provided downstream within the IOT, and RF fields are thereby induced into a cavity coupled to the gap. The RF fields may then be extracted from the output cavity in the form of a high-power, modulated RF signal. 
         [0006]    More particularly, an IOT, as well as other emission-gated microwave amplifiers, use density modulation to establish an AC current J b  on the electron beam directly at the cathode surface. This current is subsequently converted to RF energy through the J b ·E c  interaction with the output circuit field, E c . Density-modulated amplifiers are highly efficient, even when operated in the linear region. Direct modulation of the beam at the cathode also enables compact device size. 
         [0007]    In most density-modulated devices, RF gating of the electron emission is accomplished via an input cavity structure with a high-electric-field region situated between the cathode surface and a control grid. Energy from the signal generator is coupled into the input circuit, modulating the electron beam at the grid-to-cathode (g-k) gap. The basic elements of the input circuit are a resonant cavity, a coupled transmission line and a DC block. The gain-bandwidth product is limited by the interaction impedance R/Q·Q, where R/Q is the shunt impedance across the g-k gap, primarily determined by the gap geometry, and Q is the quality factor. The Q, proportional to the ratio of stored energy to dissipated power, determines the bandwidth of interaction between the drive signal and the electron beam. The power is dissipated by cavity ohmic losses (represented by Q 0 ), beam loading (Q b ) and the generator loading (Q ext ). The total Q is the parallel combination of Q 0 , Q b  and Q ext . When heavily loaded by the generator impedance through the transmission line, the cavity is strongly coupled and has a correspondingly low external quality factor (Q ext ). This reduces the total Q, which increases the bandwidth. 
         [0008]    The input resonant cavity can be modeled as a parallel RLC circuit. The beam is included as a shunt impedance and the connection to the drive line is represented by a transformer with a turns ratio of N. The Q ext  is related to the turns ratio by: 
         [0000]        N   2   Z   0   =R/Q·Q   ext , 
         [0000]    where Z 0  is the characteristic impedance of the input transmission line. Driven at its resonant frequency ω 0 , the cavity presents a purely resistive load of magnitude R=R/Q·Q to the signal generator. As the drive frequency is shifted away from ω 0 , the load becomes increasingly reactive, and the resistive component decreases. At a small offset Δω from the center frequency, the load impedance is given by: 
         [0000]    
       
         
           
             
               Z 
               load 
             
             = 
             
               R 
               
                 1 
                 + 
                 
                   2 
                    
                   j 
                    
                   
                       
                   
                    
                   Q 
                    
                   
                       
                   
                    
                   
                     Δω 
                     / 
                     
                       ω 
                       0 
                     
                   
                 
               
             
           
         
       
     
         [0009]    When the real component of the load impedance has dropped to half of its value at resonance, or R/2, the power delivered by the generator will be halved. This occurs when Δω/ω 0 =1/(2Q). Hence, the fractional bandwidth of a resonant cavity, defined as the distance between the two half-power points divided by ω 0 , is given by the reciprocal of the total quality factor (1/Q). 
         [0010]    The coupling transformer connecting the signal generator to the resonant cavity is typically implemented using an inductive loop to transfer power from the signal generator to the cavity. The degree of coupling is proportional to the ratio of the magnetic flux enclosed by the inductive loop to the total flux in the cavity. A resonant cavity is formed around the electron gun in the IOT, with the g-k gap supporting the electric fields that modulate the electron beam. The electron beam passing through the grid is bunched at the frequency of the input signal. Electrons are accelerated towards a positively biased anode before their energy is extracted by the output circuit. For existing IOT applications, such as UHF television broadcast, loop coupling provides adequate bandwidth of a few percent. Practical limits on the loop size prevent substantially larger bandwidths from being achieved. Hence, if a wide-bandwidth IOT were possible, the compactness and linearity of this device would make it an attractive option for many other applications. 
         [0011]    Accordingly, it is highly desirable to improve the instantaneous bandwidth of the input circuit of an IOT or other density-modulated device. 
       SUMMARY OF THE INVENTION 
       [0012]    The instantaneous bandwidth achievable in an IOT or other density-modulated device is increased by employing an input circuit that directly couples the radio frequency signal carried by an input coaxial transmission line to the control grid. Such a directly coupled system comprises a coaxial transmission line with one conductor connected directly to the cathode and the other connected directly to the control grid, DC isolation being provided by an appropriately located DC block. Intermediate coupling methods, such as inductive loops or capacitive probes, are not used. Several methods exist for implementing the directly coupled system. One class of implementations utilizes a resonant cavity to generate a voltage between the cathode and the control grid. In its most basic topology, the center conductor of the transmission line is connected to the cathode, while the outer conductor of the transmission line is connected to the outside wall of the resonant cavity, the outside wall also serving to support the control grid and to provide an electrical connection between the outer conductor and the control grid. In another topology employing a resonant cavity, the cathode takes the form of an annular ring supported by an annular cathode support structure within the resonant cavity. The outer conductor of the coaxial transmission line is connected to the cathode support structure. The center conductor of the coaxial transmission line extends through the center of the resonant cavity and connects to the top of the cavity, which also serves as a grid support structure, holding an annular control grid in place in close proximity to the cathode and providing an electrical connection between the grid and the center conductor of the transmission line. 
         [0013]    In both of these topologies, the impedance mismatch between the coaxial transmission line and the resonant cavity can be tuned by employing several techniques. First, an iris can be positioned at the location where the outer conductor of the coaxial transmission line joins the resonant cavity. The iris has an opening with a diameter that is smaller than that of the outer conductor of the transmission line but larger than the diameter of the center conductor, allowing the center conductor to pass through the iris. The effect of the iris is to change the magnitude of the capacitive discontinuity that appears at the transition from the coaxial transmission line to the resonant cavity. Second, various transmission line filters, well known to those skilled in the art, may be employed to change the impedance of the coaxial transmission line. For example, a slug tuner, or a parallel- or series-connected coaxial filter, such as a quarter-wave tuning stub, may be employed on the coaxial transmission line. 
         [0014]    Another class of implementations support a voltage between the cathode and the control grid without the use of a resonant cavity. In this class of implementations, the electric field propagating in the coaxial transmission line directly generates a time-varying voltage across the grid-to-cathode gap. In one non-resonant topology, the cathode is connected to the center conductor of the coaxial transmission line while the grid is connected to the outer conductor in such a way that it is positioned in close proximity to the cathode. The center conductor may terminate in a right circular cylinder, or may be shaped to affect the impedance of the transmission line and the position of the cathode attached to it. 
         [0015]    In another non-resonant topology, the cathode is connected to the outer conductor of the transmission line while the grid connects to the center conductor. To implement this, the coaxial transmission line transitions to a radial transmission line and the cathode takes the form of an annular ring connected to the bottom conductor of the radial transmission line. The control grid also takes on an annular form and is supported by the upper conductor of the radial transmission line, which also provides an electrical connection to the center conductor of the coaxial transmission line. 
         [0016]    In both of these topologies, the impedance of the coaxial transmission line can be tuned by employing slug tuners or coaxial transmission line filters as described above. Furthermore, the transmission line can be terminated by the electron beam alone or in combination with a resistive termination disposed between the cathode and the control grid. 
         [0017]    A more complete understanding of the directly coupled system providing increased operating bandwidth to IOTs and other density-modulated electron beam devices will be afforded to those skilled in the art, as well as a realization of additional advantages and objects thereof, by consideration of the following detailed description of the preferred embodiment. Reference will be made to the appended sheets of drawings which will first be described briefly. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0018]      FIG. 1  is a perspective schematic drawing of a conventional IOT; 
           [0019]      FIG. 2  is a parallel RLC circuit model, characterizing a conventional input transmission line coupled to a resonant cavity with beam impedance Z b ; 
           [0020]      FIG. 3  is a schematic layout of a conventional loop coupling in an IOT input circuit; 
           [0021]      FIG. 4(   a ) is a perspective drawing of a direct coupling system in an IOT input circuit in accordance with an embodiment of the invention, and  FIG. 4(   b ) is a cross-sectional view of a directly coupled input circuit; 
           [0022]      FIG. 5  is a parallel RLC circuit model of an input transmission line coupled to a resonant cavity, including the discontinuity capacitance, C d , that accounts for the higher-order modes arising from the change in transmission line radius; 
           [0023]      FIG. 6(   a ) is a perspective drawing of an alternative embodiment of the direct coupling system, and  FIG. 6(   b ) is a cross-sectional view of this alternative embodiment; 
           [0024]      FIGS. 7(   a )-( f ) are alternative embodiments of the non-resonant direct coupling system terminated in the beam impedance; and 
           [0025]      FIGS. 8(   a )-( f ) are alternative embodiments of the non-resonant direct coupling system terminated in a resistive load. 
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
       [0026]    The invention provides improved instantaneous bandwidth of the input circuit of an IOT or other density-modulated device. In the detailed description that follows, like numbers are used to describe like elements illustrated in one or more of the figures. 
         [0027]      FIG. 1  is a schematic drawing of an exemplary IOT, typical of the prior art. The IOT includes three major sections, including an electron gun  150 , a tube body  160 , and a collector  170 . The electron gun  150 , shown in more detail in  FIG. 3 , provides an axially directed electron beam that is density modulated by an RF signal. The electron beam passes through a first drift tube  230  and a second drift tube  232  and then passes into an inner structure  234  inside the collector  170  that collects the spent electron beam. The electron gun further includes a cathode  206  with a closely spaced control grid  204 . The cathode is disposed at the end of a cylindrical capsule  208  that includes an internal heater coil  220  coupled to a heater voltage source  224 . The control grid  204  is positioned closely adjacent to the surface of the cathode  206 , and is coupled to a bias voltage source to maintain a DC bias voltage relative to the cathode. A resonant input cavity  202  receives an RF input signal via a coaxial transmission line  210 . The RF signal is coupled between the control grid  204  and cathode  206  to density modulate the electron beam emitted from the cathode  206 . The control grid is physically held in place by a grid support structure  226 . An example of an input cavity for an inductive output tube is provided by U.S. Pat. No. 6,133,786, the subject matter of which is incorporated in its entirety by reference herein. 
         [0028]      FIG. 2  depicts a parallel RLC circuit model of a conventional input circuit of the prior art. The electron beam is modeled as a shunt impedance  112  of Z b , and the resonant cavity is modeled as a parallel combination of a resistor  106 , an inductor  108 , and a capacitor  110 . The input transmission line  102 , with a characteristic impedance of Z 0 , is coupled to the resonant cavity via an inductive loop, modeled as a transformer  104 , with an effective turns ratio of N. As discussed previously, this results in a load impedance presented to the input transmission line due to the cavity of 
         [0000]    
       
         
           
             
               
                 Z 
                 load 
               
               = 
               
                 R 
                 
                   1 
                   + 
                   
                     2 
                      
                     j 
                      
                     
                         
                     
                      
                     Q 
                      
                     
                         
                     
                      
                     
                       Δω 
                       / 
                       
                         ω 
                         0 
                       
                     
                   
                 
               
             
             , 
           
         
       
     
         [0000]    where Δω represents a small offset from the cavity resonant frequency ω 0 . Using this expression to calculate the half power points, the fractional bandwidth of the system is obtained as 1/Q, where Q is the quality factor. 
         [0029]      FIG. 3  represents an exemplary physical layout of the conventional prior-art input circuit modeled in  FIG. 2 . The coupling transformer is implemented as an inductive loop  212  that couples energy from the input coaxial transmission line  210  into the resonant cavity  202 . The cathode  206  is situated atop a cathode support structure  208  to place it in close proximity to a control grid  204  that permits passage of the electron beam emitted by the cathode  206 . The cavity geometry places practical limitations on loop size, and as a consequence, limits the fraction of the magnetic flux that is intercepted, restricting this technique to applications requiring relatively narrow bandwidths. 
         [0030]    The invention described herein discloses a method for coupling to the input circuit of an IOT or other emission-gated device that allows for a substantially lower Q ext  that is able to achieve substantially greater bandwidths. This is achieved by providing a coaxial transmission line that directly couples to the cavity surrounding the grid-to-cathode interaction region. This direct coupling results in a relatively low external quality factor (Q ext ) that reduces the total Q, increasing the bandwidth of the input circuit. 
         [0031]    Several implementations of the directly coupled input circuit are possible. The most basic embodiment of the invention is shown in  FIGS. 4(   a ) and  4 ( b ).  FIG. 4(   a ) presents a three-dimensional view of the input circuit, and  FIG. 4(   b ) presents an axial cross-sectional view of the input circuit. Like numbers are used to refer to corresponding structures between the two figures. In this embodiment, the center conductor  316  of the coaxial input transmission line transitions to the cathode support structure  312 , and the outer conductor  318  is connected to the outside wall of the cavity  308 . A control grid  306  is connected to the wall of the cavity  308  and held in close proximity to the cathode  310 , which is situated at the top of the cathode support structure  312 . A DC block is located between the outer conductor  318  and the grid  306  to enable a DC bias to be maintained between the grid  306  and the cathode  310  while permitting direct coupling of the RF signal from the transmission line to the grid. An optional iris  314 , in the form of an annular ring, may be disposed at the location where the outer conductor  318  of the transmission line joins the cavity wall  308 . The diameter of the opening of the iris  314  is larger than the diameter of the cathode support structure  312 , but smaller than the diameter of the outer conductor  318 . In the discussion that follows, the diameter of the iris opening  322  is represented by 2r a . The diameter of the resonant cavity  320  is represented by 2r c . The inner diameter of the outer conductor  324  and the diameter of the center (inner) conductor  326  of the transmission line are represented by 2r o  and 2r i , respectively. Though  FIGS. 4(   a ) and  4 ( b ) depict a center conductor that is a right circular cylinder in shape, the center conductor may be stepped or tapered, such as the center conductor depicted in  FIG. 7(   c ), in order to modify the impedance of the coaxial transmission line. 
         [0032]    The geometry represented in  FIGS. 4(   a ) and  4 ( b ) can be modeled by the equivalent circuit shown in  FIG. 5 . The beam impedance, Z b , is modeled as a shunt  412 . The cavity is modeled as a parallel RLC circuit including a resistor  406 , an inductor  408 , and a capacitor  410 . The coupling of the coaxial transmission line  402  to the cavity is modeled as a transformer  404  as well as a shunt capacitance  414 , called the discontinuity capacitance, C d , to account for the higher order modes excited at the impedance step that results from the change in diameter as a signal leaves the coaxial transmission line and enters the resonant cavity. The turns ratio of the transformer, N, is approximately 
         [0000]        N   2   ≈Z   cp   /Z   tl . 
       The cavity port impedance, Z cp , and the transmission line impedance, Z tl , are given by 
       [0033]        Z   cp =[(μ/∈) 1/2 /2 π]In ( r   c   /r   i ), and 
         [0000]        Z   tl =[(μ/∈) 1/2 /2 π]In ( r   0   /r   i ), 
         [0000]    where r 0  and r c  are the radii of the outer conductor  318  of the coaxial transmission line and the resonant cavity  308  respectively, and r i  is the radius of the center (inner) conductor  316 . The calculation of the discontinuity capacitance, C d , requires a full field solution. The Q ext  of the cavity is defined as Q ext =ω 0 U/P l , where U is the energy stored in the cavity and P l  is the power dissipated in the transmission line load. 
         [0034]    This power, defined as P l =½l 2 R, requires calculation of the current, l, flowing out of the cavity into the transmission line. The shunt capacitance in parallel with this load acts as a current divider. The fraction of the current that flows through the transmission line load is 1/(α 2 +1), where α=N 2 Z tl ω 0 C d . Since Q is inversely proportional to l 2 , the reduction in current modifies the Q ext  defined above, resulting in: 
         [0000]    
       
         
           
             
               Q 
               ext 
             
             = 
             
               
                 
                   
                     N 
                     2 
                   
                    
                   
                     Z 
                     tl 
                   
                 
                 
                   R 
                   / 
                   Q 
                 
               
                
               
                 
                   ( 
                   
                     
                       α 
                       2 
                     
                     + 
                     1 
                   
                   ) 
                 
                 2 
               
             
           
         
       
     
         [0035]    For a typical design at L-band, the discontinuity capacitance is on the order of 0.1 picofarads, resulting in α≈0.1, and hence Q ext ≈Z cp /R/Q. Depending on the specific geometry, very low Q ext , approaching unity, can be achieved. 
         [0036]    If an iris  314  is included, where r a &lt;r 0 , the discontinuity capacitance is increased, shunting a larger portion of the current and increasing the Q ext  without changing the cavity or transmission line geometry. A tapered or stepped transmission line or other impedance transformer may be used in place of, or in conjunction with, the iris to change the transmission line impedance presented to the cavity. Placement of a filter network in the transmission line offers further control of the bandwidth. An example of this, well known to those skilled in the art, is a coaxial impedance transformer, such as a slug tuner, on the center of the transmission line.  FIG. 4(   a ) depicts a dielectric slug tuner  320  used to tune the impedance of the input line. Another example of such a filter network is a transmission line resonant cavity, connected either in series or in parallel, such as the tuning stub  520  depicted in  FIG. 6(   b ). 
         [0037]      FIGS. 6(   a ) and  6 ( b ) illustrate a second embodiment of the direct coupling system. A three dimensional view is depicted in  FIG. 6(   a ), and a cross-sectional view is presented in  FIG. 6(   b ). Like numbers are used to refer to corresponding structures. A ring cathode  510  is mounted on an annular support structure  512 , and this support structure is connected to the outer conductor  518  of the transmission line. The center conductor  514  of the transmission line extends through the cavity and is connected to a grid support structure  520  that supports an annular control grid  506  and further provides an electrical connection between the center conductor  514  and the control grid  506 . A DC block is located between the outer conductor  518  and the grid  506  to enable a DC bias to be maintained between the grid  506  and the cathode  510  while permitting direct coupling of the RF signal from the transmission line to the grid. An optional iris  516  may be used to alter the magnitude of the discontinuity capacitance between the coaxial transmission line and the cavity  508 . An optional stub tuner  520 , may likewise be used to tune the impedance of the coaxial transmission line to alter the magnitude of the discontinuity capacitance. Using a coaxial impedance transformer, a cold test model of this embodiment has been fabricated and tested, and has achieved an instantaneous bandwidth in excess of twenty percent. 
         [0038]    The voltage across the grid-to-cathode gap need not be provided by a resonant cavity. Instead, the electric field of the transmission line mode may be used to generate the voltage in a non-resonant directly coupled system. A portion of the power carried by the transmission line is coupled into the electron beam. Termination of the transmission line in its characteristic impedance results in maximum bandwidth. The termination can be provided by the beam as illustrated in  FIGS. 7(   a )- 7 ( f ), by a resistive load located after the beam as illustrated in  FIGS. 8(   a )- 8 ( f ), or by some combination of the two. A transmission line transformer, such as a slug tuner or resonant cavity filter, may be used to facilitate the match. 
         [0039]      FIGS. 7(   a )-( f ) show three possible embodiments of the non-resonant direct coupling system.  FIG. 7(   a ) represents a three-dimensional view and  FIG. 7(   b ) represents a cross-sectional view of a cylindrical non-resonant directly coupled system. The cathode  608  is disposed at the end of the center conductor  610  of the input coaxial transmission line. The outer conductor  612  of the transmission line is connected to the control grid  606 . The voltage across the grid-to-cathode gap, between the cathode  608  and grid  606 , is provided by the electric field of the electromagnetic wave traveling in the coaxial transmission line. The termination of the transmission line is provided by the electron beam itself. 
         [0040]      FIGS. 7(   c ) and  7 ( d ) show an alternative embodiment in which the center conductor  628  is tapered. The cathode  626  surrounds the tapered end of the center conductor  628  and is held in close proximity to the control grid  624  that is situated around the tapered center conductor. The outer conductor  630  is connected to the control grid  624 . Varying the geometry of the tapered center conductor will change the impedance of the transmission line, which is terminated by the electron beam itself. 
         [0041]      FIGS. 7(   e ) and  7 ( f ) depict an alternative embodiment of the non-resonant directly coupled system. In this embodiment, the coaxial transmission line comprising a center conductor  658  and an outer conductor  660 , transitions to a radial transmission line. The center conductor  658  attaches to the annular control grid  654 . The annular cathode  656  is attached to the lower wall of the radial transmission line and connected directly to the outer conductor  660  of the coaxial transmission line. In this embodiment, as well, the transmission line is terminated by the electron beam. 
         [0042]      FIGS. 8(   a )-( f ) present the same embodiments of the non-resonant direct coupling system shown in  FIGS. 7(   a )-( f ), except that here the termination is provided by a resistive load rather than solely by the electron beam. In  FIGS. 8(   a ) and  8 ( b ), the resistive load  714  is situated between the cathode  708  and the control grid  706 . Similarly in  FIGS. 8(   c ) and  8 ( d ), the resistive load  732  is placed between the center conductor  728  and the control grid  724  that is connected to the outer conductor  730 . Finally, in  FIGS. 8(   e ) and  8 ( f ), the resistive load  762  is situated around the outside of the radial transmission line between the cathode  756 , connected to the outer conductor  760 , and the grid  754 , connected to the center conductor  758 . It should be noted that the beam can be emitted from a cathode connected either to the center conductor, as shown in  FIGS. 7(   a )-( d ) and  8 ( a )-( d ), or to the outer conductor, as shown in  FIGS. 7(   e )-( f ) and  8 ( e )-( f ). 
         [0043]    It should be appreciated that the above-described geometries are not meant to be comprehensive but are representative embodiments of the present invention that utilize direct coupling of a transmission line to achieve wideband coupling from the transmission line to the electron beam. By employing the direct coupling system, this invention enables inductive output devices to be adapted for service in wide-instantaneous-bandwidth applications. The method is also likely to spur the development of other novel emission-gated devices, employing thermionic and non-thermionic cathodes. 
         [0044]    Having thus described a preferred embodiment of a novel input circuit that provides improved instantaneous bandwidth for an inductive output tube or other emission-gated device, it is apparent to those skilled in the art that certain advantages of such systems have been achieved. It should also be appreciated that various modifications, adaptations, and alternative embodiments thereof may be made within the scope and spirit of the present invention. The invention is further defined by the following claims.