Abstract:
A method and circuit for receiving a radio frequency signal by receiving and amplifying the radio frequency signal to produce a received signal and generating first and second clock signals corresponding to first and second channel signals, respectively, of the received signal and multiplying the received signal with the clock signals to obtain the channel signals. Pre-selectivity filtering of the received signal is performed by filtering the first channel using a first impedance, filtering the second channel using a second impedance, and converting the first and second impedances with respect to one another through a first gyrator. Amplitude limiting of the first and second channels is performed to obtain first and second amplitude limited channels. Poly-phase selectivity filtering of the first and second amplitude limited channels is performed to obtain first and second selectivity filtered channels. The selectivity filtered channels are demodulated to obtain a data signal.

Description:
CROSS-REFERENCE TO RELATED APPLICATION 
     This application claims priority to U.S. provisional application No. 60/668,637 entitled CIRCUIT AND METHOD FOR SIGNAL RECEPTION USING A LOW INTERMEDIATE FREQUENCY filed Apr. 6, 2005, and to U.S. provisional application No. 60/668,922 entitled DIFFERENTIAL SLOPE FM DEMODULATOR FOR LOW-IF FREQUENCIES filed Apr. 6, 2005, herein incorporated by reference in their entirety for all purposes. This application is related to commonly owned U.S. provisional application No. 60/668,922 entitled DIFFERENTIAL SLOPE FM DEMODULATOR FOR LOW-IF FREQUENCIES filed Apr. 6, 2005 and corresponding U.S. patent application Ser. No. 11/399,058, now U.S. Pat. No. 7,405,613, herein incorporated by reference in their entirety for all purposes. 
    
    
     FIELD OF INVENTION 
     The present invention relates to reception of a data signal and, more specifically, to signal reception using an intermediate frequency (IF). 
     BACKGROUND OF THE INVENTION 
     The cost and size of a single chip receiver can be reduced when the receiver&#39;s selectivity filter and the demodulator are completely integrated on the chip. To obtain reasonable power consumption, which is required for longer battery life, the intermediate frequency (IF) chosen for the receiver is typically relatively low, e.g. 200 kHz. 
     In general, the image reception is suppressed by a combination of a quadrature mixer and a poly-phase filter. The low IF outputs (the I and Q channel) of the quadrature mixer are connected to the input of a poly phase selectivity filter that inherently suppresses the image frequency. The first stages of the poly phase filters are soft-clipping amplifiers to prevent non linear distortion, such as AM to PM conversion, caused by large signals that exceed the linear range of the integrated poly phase filter. Alternatively, the soft-clipping amplifier may be replaced with an automatic gain controlled front-end. Examples of such architectures are found in Brian J. Minnis and Paul A. Moore, “Improvements in or relating to poly phase receivers,” PCT application no. WO 01/39385 A1, May 31, 2001 (“Minnis et al.”), and H. van Rumpt, W. G. Kasperkovitz and J van der Tang, “A digitally programmable zero external components FM radio receiver with 1 uV sensitivity,” ISSCC 2003 (“van Rumpt, et al.”). In a circuit implementation, it is common practice to implement a pre-selectivity filter (e.g. a first order, low-pass filter) immediately following the mixer outputs to create some selectivity before the signal enters the poly phase selectivity filter. 
     The cut-off frequency of these pre-selectivity filters needs to be at least IF+(Bw/2), where Bw is the signal bandwidth. On the image side, however, these low-pass filters will not reduce the interference up to 2*IF+(Bw/2) frequency distance from the receive channel. The dynamic range of the poly phase filters needs to be high enough to cope with the unfiltered interference at the image side. 
     SUMMARY OF THE INVENTION 
     In one embodiment of a low intermediate frequency receiver, according to one aspect of the present invention, the receiver includes an amplifier having an input for receiving a radio frequency signal and an output for outputting the received signal. The receiver also includes a local oscillator having a first output for providing a first clock signal corresponding to a first channel of the received signal and a second output for providing second clock signal corresponding to a second channel of the received signal. A quadrature mixer has an input coupled to the output of the amplifier, a first clock input coupled to the first output of the local oscillator, and a second clock input coupled to the second output of the local oscillator, such that the received signal is input to the quadrature mixer along with the first and second clock signals and, responsive thereto, the quadrature mixer outputs the first channel signal at a first output and the second channel signal at a second output. The first and second outputs of the quadrature mixer are coupled to a pre-selectivity filter configured to remove an image channel from the first and second channel signals. The first output of the quadrature mixer is further coupled to a first amplitude limiter circuit and the second output of the quadrature mixer is further coupled to a second amplitude limiter circuit. Outputs of the first and amplitude limiter circuits are coupled to inputs of a selectivity filter circuit. An output of the selectivity filter circuit is coupled to a demodulator. The pre-selectivity filter further comprises a first resistive-capacitor network coupled to the first output of the quadrature mixer and a second resistive-capacitor network coupled to the second output of the quadrature mixer, where the first and second resistive-capacitor networks are coupled together through a gyrator. In a further refinement of this embodiment, the selectivity filter is a poly-phase selectivity filter further that includes gyrator resonator type poly-phase filter elements. In still another refinement of this embodiment, the poly-phase selectivity filter uses the same type of devices as the gyrator resonator and the amplitude limiter further comprise soft clippers that include transconductors that are the same type of device and topology as is used in the quadrature mixer. 
     In an embodiment of a method, according to the present invention, for receiving a radio frequency signal, the radio frequency signal is received and amplified to produce a received signal. First and second clock signals are generated, where the first clock signal corresponds to a first channel of the received signal and the second clock signal corresponds to a second channel of the received signal. The received signal is mixed with the first and second clock signals to obtain the first and second channel signals. The first channel signal is pre-selectively filtered using a first impedance and the second channel signal is pre-selectively filtered using a second impedance. The first and second channel signals are then amplitude limited and the amplitude limited first and second channel signals are selectively filtered. The selectively filtered first and second channel signals are then demodulated to obtain a data signal. The first and second impedances are converted with respect to one another through the use of a gyrator effect. 
    
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
       Certain embodiments of the present invention are described below with reference to the following drawings wherein: 
         FIG. 1  is a functional block diagram illustrating an example of a conventional low IF receiver circuit; 
         FIG. 2  is a frequency response diagram illustrating the frequency response of the mixer and pre-selectivity filter of  FIG. 1 ; 
         FIG. 3  is a functional block diagram illustrating an embodiment of a circuit having pre-selectivity filtering involving a gyrator; 
         FIG. 4  is a frequency response diagram illustrating an example of a frequency response of the mixer and pre-selectivity filter circuit of  FIG. 3 ; 
         FIG. 5  is a functional block diagram illustrating an embodiment of a circuit having a matched gyrator selectivity filter; 
         FIG. 6  is a functional block diagram illustrating an example of a differential gyrator circuit suitable for use in the circuits of  FIGS. 3 and 5 ; and 
         FIG. 7  is a functional block diagram illustrating another embodiment of a poly phase pre-selectivity filter. 
     
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS 
     A method and circuit for low intermediate frequency reception is shown that utilizes gyrator resonator pre-selectivity filtering of a received image so that subsequent elements may operate with reduced dynamic range. In a different aspect of the present invention, gyrator resonator type poly-phase filter elements are used in the receiver to obtain accurate and substantially temperature independent matching of center frequency and bandwidth. 
     The invention is based on the insight that the pre-selectivity filtering can be much more effective when poly phase filtering is merged into a quadrature mixer implementation in a manner described below. One aspect of the present invention is to provide filtering in a more effective way so that the dynamic range requirements of the poly phase IF filter can be relaxed, which may permit power consumption and chip area to be reduced. 
     An example of a common implementation of a low-IF receiver, such as those taught by Minnis et al. and van Rumpt et al., is sketched in  FIG. 1 . In this example, the signal connections are drawn as differential signals. 
       FIG. 1  is a functional block diagram illustrating an example of a conventional low IF receiver circuit  10 , such as those described by Minnis, et al. and van Rumpt, et al. A low noise amplifier  12  receives a radio frequency (RF) signal via an antenna  14 . Amplifier  12  outputs a differential RF signal that is input to a quadrature mixer circuit  20 , where the RF signal is input to multipliers  28  and  30  to obtain an I channel and a Q channel from the received signal. Local oscillator  16  provides an I channel local oscillator signal LO I , which is input to multiplier  30 , and a Q channel local oscillator signal LO Q , which is input to multiplier  28 . The local oscillator  16  is tuned to the desired channel frequency plus or minus the Intermediate Frequency (IF) and produces the in-phase (LO I ) and quadrature-phase (LO Q ) outputs. The output of multiplier  28  is subject to pre-selectivity filtering by the combination of resistors  22  and  26  with capacitor  24 . Likewise, the output of mixer  30  is subject to pre-selectivity filtering by the combination of resistors  32  and  36  with capacitor  34 . The resulting I and Q channels are input to soft clippers  40  and  42 , which are amplitude limiters that limit the maximum amplitude of the channel signals, and then input to poly-phase selectivity filter  50 . The output of poly-phase selectivity filter  50  is input to demodulator  60 , which outputs the received data signal. 
     Most multipliers have current outputs, e.g. a Gilbert multiplier is commonly used. See “A Precise Four Quadrant Multiplier with Subnanosecond Response,” Barrie Gilbert, IEEE Journal of Solid State Circuits, Vol. SC-3, No. 4, December 1968.This current output is transformed to a voltage and high frequency components are filtered by the R psf -C psf  combinations. In this example, this filter is called a “pre-selectivity filter”. 
     The result of this mixing process is that the RF signal is transposed to an IF signal obtained at the output of the Quadrature Mixer  20  in both the in-phase I channel and quadrature-phase Q channel. The limitation of the typical prior art pre-selectivity filtering approach is illustrated in the frequency response diagram of  FIG. 2 .  FIG. 2  is a frequency response diagram illustrating the filer limitation of the conventional pre-selectivity filtering of  FIG. 1 . In  FIG. 2 , the local oscillator frequency is F LO , the desired channel appears at F LO +IF, and the image channel appears at F LO −IF. 
     In the conventional approach, no image suppression occurs at the output of the mixer  20 . Adjacent channels that are present at the image site are passed right through to the soft clippers without any filtering up to a frequency distance (with respect to the center of the wanted channel) of 2*IF+(Bw/2). Consequently, the dynamic range of the successive soft clippers and poly-phase filters needs to be sufficiently high to filter the image. Because the image is still present, greater dynamic range is required in order to filter the image, which results in a circuit that requires greater supply current. 
     Another potential problem that may be associated with the conventional approach is that this asymmetrical filtering leads to an asymmetrical receive channel. The succeeding circuitry needs to be adapted to accommodate the asymmetrical channel, which leads to sub-optimal receiver design. Examples of potential problems caused by asymmetry in the receive channel are generation of second order distortion products and an increase of inter-symbol interference. 
     Still another potential problem that may be associated with the conventional approach is that the adjacent selectivity performance on the image side completely depends on the poly-phase selectivity filter  50 . The pre-selectivity filter within quadrature mixer  20  has no selectivity contribution in the adjacent channels at the image side. 
     Yet another potential problem that may be associated with the conventional approach is that the IF signal has to pass through the soft clippers  40  and  42  before it enters the poly-phase selectivity filter  50 . Due to phase and amplitude mismatch introduced by the soft clippers  40  and  42  and the associated interconnection to poly-phase selectivity filter  50 , the I Q relationship is distorted, which will result in a degradation of image suppression. 
     One aspect of the present invention is to provide filtering in a more effective way so that the dynamic range requirements of the poly-phase IF filter can be relaxed. Power consumption and chip area may also be reduced. 
     A second aspect of the present invention is to obtain a receive channel that is completely symmetrical. This allows for a better design for the overall channel selectivity that potentially improves the receiver in terms of selectivity and distortion. At the same time, the receiver complexity can be reduced, since the impairments caused by asymmetry no longer require compensation provisions. 
     A third aspect of the present invention is to improve the intrinsic image suppression without the use of complex compensation loops. 
     The present invention is based on the insight that the pre-selectivity filtering can be much more effective when poly phase filtering is merged into the quadrature mixer implementation in the manner described below. In accordance with the present invention, by adding a gyrator resonator to the output of the mixers and pre-selectivity filter, the image channel can be suppressed before it is input to the soft clippers and selectivity filter of the receiver. Thus, the dynamic range of the selectivity filter may be reduced, which can lead to less current draw and reduced chip area. 
       FIG. 3  is a circuit diagram illustrating one exemplary embodiment of a receiver circuit  100  according to the present invention. In the circuit of  FIG. 3 , a gyrator circuit  122  is added to the quadrature mixer  120 . See “Integration of Analog Filters in a Bipolar Process,” J. O. Voorman, W. H. A. Brüls and P. J. Barth, IEEE Journal of Solid-State Circuits, Vol. SC-17, No. 4, August, 1982, for one example of a gyrator circuit. A gyrator is basically an impedance converter and is formed essentially by two anti-parallel arranged transconductors, one of which has a positive transconductance G and the other a negative transconductance −G. 
     In receiver  100 , the preselectivity filter networks (e.g. resistors  22  and  26  with capacitor  24 , resistors  32  and  34  along with capacitor  36 ) implement quadrature R psf -C psf  networks that work together with the gyrator action of gyrator  122 , e.g. forming a gyrator resonator, to effectuate poly-phase filtering of the low-IF signal even before the low-IF signal enters the soft clippers  40  and  42 . In this embodiment, active elements within the gyrator resonator  122  are benefiting to the same extent as the soft clippers and the poly-phase filters. By dimensioning the passive elements values R psf , C psf  and the gyrator constant G psf , the filter center frequency and bandwidth can be set to fit the chosen low-IF receiver specification. Note that a first order filter is shown. Higher order filters may be implemented by adding one gyrator resonator and two resistive capacitor networks for each filter. For example, a fifth order filter would typically involve five gyrator resonators and ten RC networks. 
     Due to the improved filtering of the present invention, the dynamic range of the succeeding circuits, like the soft clippers and the poly-phase selectivity filter, can be relaxed. Power consumption and chip area can, therefore, be reduced.  FIG. 4  is a frequency response diagram that illustrates the effectiveness of the poly-phase pre-selectivity filtering of the receiver  100  of  FIG. 3 . Since the frequency response of the poly-phase pre-selectivity filter is symmetrical around the wanted channel, it fully contributes to the receiver&#39;s channel selectivity. This means that one filter order can be omitted in the requirements for the design of the poly-phase selectivity filter, which, again, can result in an additional reduction of both power consumption and chip area. Furthermore, the resulting receive channel is substantially completely symmetrical allowing for improved overall channel selectivity filter design, reduced complexity and boosting receiver performance, while reducing second order distortion and inter-symbol interference. 
       FIG. 5  is a circuit diagram illustrating another embodiment of a low-IF receiver  200 , wherein the poly-phase selectivity filter  250  is constructed with a gyrator resonator type poly-phase filter element  252 . See Voorman et al. and U.S. Pat. No. 4,193,033 for examples of gyrator resonator elements, as well as  FIG. 6 . Device and layout matching is used between the poly-phase pre-selection filter  120  and the poly-phase selectivity filter  250  to obtain accurate and substantially temperature independent matching of center frequency and bandwidth. 
     In the embodiment of  FIG. 5 , the poly-phase selectivity filter  250  uses the same type of devices as the poly-phase pre-selectivity filter  120  and the soft clipping transconductors  240  and  242  use the same type of devices and topology as is used in the multipliers  28  and  30 , e.g. bipolar differential pairs. Capacitor  254  and resistor  256  are coupled to the output of soft clipping transconductor  240  and implement a selectivity filter based on R f C f  for the I channel. Capacitor  264  and resistor  266  are coupled to the output of soft clipping transconductor  242  and implement a selectivity filter based on R f C f  for the Q channel. The R f C f  networks work together with the gyrator action of gyrator resonator  252  to effectuate poly-phase selectivity filtering of the low-IF signal. In this embodiment, the soft clippers  240  and  242  benefit from gyrator  122 . The active element devices within the gyrator resonator  252  can be matched the active elements of the poly-phase pre-selectivity filter in quadrature mixer  120 . By dimensioning the passive elements values R f , C f  and the gyrator constant G f , the filter center frequency and bandwidth can be set to fit the chosen low-IF receiver specification. 
     In the circuit  200  of  FIG. 5 , only a first order poly-phase filter is shown. It will be evident to one of ordinary skill in the art that a plurality of gyrator resonator type poly phase elements can be applied to comply with substantially any selectivity filter specification without departing from the teachings of the present invention. Likewise, while the embodiments shown utilize a differential approach, the teachings of the present invention may be applicable to single-ended solutions without departing from the teachings of the present invention, as will be readily appreciated by one of ordinary skill in the art. 
       FIG. 6  shows one example of a differential gyrator resonator  300  suitable for use in the gyrator resonator circuit elements discussed above in  FIGS. 3 and 5 . In the gyrator resonator  300  of  FIG. 6 , transconductor amplifiers  302  and  306  (with transconductances G 1  and G 2 , respectively) are coupled in anti-parallel fashion. The two transconductors  302  and  306 , together with the capacitors  24  and  34 , e.g. capacitance C psf , in quadrature mixer  120 , form a gyrator resonator within the quadrature mixer  120 . The resonator is damped by the R psf  resistors  22 ,  26 ,  32  and  36  to create the desired bandwidth. At the nodes I and IB, the capacitors connected at the nodes Q and QB are seen as inductances due to the gyrator principle and, at the nodes Q and QB, the capacitors connected at the nodes I and IB are seen as inductances, hence an LC-like parallel resonator is formed. The resonance frequency is determined by G 1 , G 2 , and C psf . In the preferred embodiment, G psf =G 1 =G 2 . 
     Similarly, the two transconductors  302  and  306 , together with the capacitors  254  and  264 , e.g. capacitance C f , in poly-phase selectivity filter  250 , form a gyrator resonator within the poly-phase selectivity filter  250 . The resonator is damped by the R f  resistors  256  and  266  to create the desired bandwidth. 
     In a preferred embodiment the resonator components including its values and layout are substantially symmetrical, e.g. the components driving the nodes I and IB are symmetrical with the components driving the nodes Q and QB. Furthermore, all transconductors have substantially the same dependencies, which means that their transconductances as a function of biasing, temperature, process spread, operating voltage, etc. are essentially the same. For further details regarding gyrator resonator circuits, see U.S. Patent Application No. 60/606,037 entitled “TUNABLE POLY-PHASE FILTER AND METHOD FOR CALIBRATION THEREOF” filed Aug. 31, 2004, herein incorporated by reference in its entirety for all purposes. 
     The center frequency of the poly-phase pre-selectivity filter within quadrature mixer  120  is determined by: 
     
       
         
           
             
               
                 
                   
                     Fc 
                     psf 
                   
                   = 
                   
                     
                       G 
                       psf 
                     
                     
                       2 
                       · 
                       π 
                       · 
                       
                         C 
                         psf 
                       
                     
                   
                 
               
               
                 
                   ( 
                   1 
                   ) 
                 
               
             
           
         
       
     
     In the same way the center frequency of the poly-phase selectivity filter  250  is determined by: 
     
       
         
           
             
               
                 
                   
                     Fc 
                     f 
                   
                   = 
                   
                     
                       G 
                       f 
                     
                     
                       2 
                       · 
                       π 
                       · 
                       
                         C 
                         f 
                       
                     
                   
                 
               
               
                 
                   ( 
                   2 
                   ) 
                 
               
             
           
         
       
     
     The bandwidth of the poly-phase pre-selectivity filter within quadrature mixer  120  is determined by: 
     
       
         
           
             
               
                 
                   
                     Bw 
                     psf 
                   
                   = 
                   
                     1 
                     
                       π 
                       · 
                       
                         R 
                         psf 
                       
                       · 
                       
                         C 
                         psf 
                       
                     
                   
                 
               
               
                 
                   ( 
                   3 
                   ) 
                 
               
             
           
         
       
     
     Likewise the bandwidth of the poly-phase selectivity filter  250  is determined by: 
     
       
         
           
             
               
                 
                   
                     Bw 
                     f 
                   
                   = 
                   
                     1 
                     
                       π 
                       · 
                       
                         R 
                         f 
                       
                       · 
                       
                         C 
                         f 
                       
                     
                   
                 
               
               
                 
                   ( 
                   4 
                   ) 
                 
               
             
           
         
       
     
     The matching properties of the preferred embodiment are also apparent from these equations. 
     It will be clear that a person skilled in the art can replace the resistors (R psf  and R f ) by transconductors or transconductor-resistor combinations to create a tunable bandwidth without departing from the present invention. 
     In addition to the embodiments described above, a poly-phase demodulator  260  can be used that is matched to the pre-selectivity and the selectivity filters by using gyrator type resonators and/or oscillators. Device and layout matching is used between the poly-phase pre-selection filter in quadrature mixer  120 , the poly-phase selectivity filter  250  and a poly-phase demodulator  260  to obtain accurate and substantially temperature independent matching of center frequency and bandwidth over the complete receiver chain. 
     Examples of such poly-phase demodulators  260  includes the differential slope FM demodulator of U.S. provisional application No. 60/668,922 entitled DIFFERENTIAL SLOPE FM DEMODULATOR FOR LOW-IF FREQUENCIES filed Apr. 6, 2005, herein incorporated by reference in their entirety for all purposes. Other examples include FM and AM phased-locked-loop (PLL) demodulators, wherein the voltage controlled oscillator (VCO) is constructed by a gyrator type resonator in oscillation mode. Another example is the quadrature demodulator shown in U.S. Pat. No. 5,341,107. 
     These are only examples and other types of demodulators may also be suitable for use with the present invention. 
       FIG. 7  is a functional block diagram illustrating another embodiment of a poly phase pre-selectivity filter. In circuit  300 , the preselectivity filter networks (e.g. resistors  22 ,  26 ,  322  and  326  with capacitor  324 , resistors  32 ,  34 ,  332  and  336  along with capacitor  334 ) implement quadrature RC networks that work together with the gyrator action of gyrator  322 , e.g. forming a gyrator resonator, to effectuate poly-phase filtering of the low-IF signal even before the low-IF signal enters, for example, the soft clippers  40  and  42  of  FIG. 3 . This embodiment permits the mixer function and the pre-selectivity filtering to be determined separately. Mixers  28  and  30  are mainly optimized by adjusting the value of R psf1  for resistors  22 ,  26 ,  32  and  36 . The pre-selectivity filter is primarily set by the value of R psf2  for resistors  322 ,  326 ,  332  and  334 . By dimensioning the passive elements values R psf1 , R psf2 , and C psf  and the gyrator constant G psf , the filter center frequency and bandwidth can be set to fit the chosen low-IF receiver specification and the mixer function may be fine tuned. Note that a first order filter is shown. Higher order filters may be implemented by adding one gyrator resonator and two resistive capacitor networks for each filter. 
     Also note that, in a preferred embodiment, the gyrator resonators of the pre-selectivity, selectivity and demodulator circuits are matched devices. As a result, when one gyrator resonator is aligned, the other gyrator resonators will also be aligned and the process compensation for one device typically compensates the other devices. In order to obtain this scenario the gyrator circuits and associated components, such as the RC networks, are matched. 
     All references, including publications, patent applications, and patents, cited herein are hereby incorporated by reference to the same extent as if each reference were individually and specifically indicated to be incorporated by reference and were set forth in its entirety herein. 
     The use of the terms “a” and “an” and “the” and similar referents in the context of describing the invention (especially in the context of the following claims) are to be construed to cover both the singular and the plural, unless otherwise indicated herein or clearly contradicted by context. Recitation of ranges of values herein are merely intended to serve as a shorthand method of referring individually to each separate value falling within the range, unless otherwise indicated herein, and each separate value is incorporated into the specification as if it were individually recited herein. All methods described herein can be performed in any suitable order unless otherwise indicated herein or otherwise clearly contradicted by context. The use of any and all examples, or exemplary language (e.g., “such as”) provided herein, is intended merely to better illuminate the invention and does not pose a limitation on the scope of the invention unless otherwise claimed. No language in the specification should be construed as indicating any non-claimed element as essential to the practice of the invention. 
     Preferred embodiments of this invention are described herein, including the best mode known to the inventors for carrying out the invention. It should be understood that the illustrated embodiments are exemplary only, and should not be taken as limiting the scope of the invention.