DIGITAL TO ANALOG CONVERSION METHOD AND APPARATUS

Abstract: A digital-to-analog circuit arrangement that can be used, for example, in radio transmitters, includes a Digital-to-Analog Converter (DAC) whose output is connected directly to a mixer. A periodic signal is also provided to the mixer. The frequency of the periodic signal provided to the mixer, is an integer multiple of half the update rate frequency of the DAC. In an embodiment of the invention, the frequency of the periodic signal provided to the mixer, is half the update rate frequency of the DAC. An intermediate frequency (IF) filter can be provided downstream of the mixer.

Description: DIGITAL TO ANALOG CONVERSION METHOD AND APPARATUS BACKGROUND OF THE INVENTION 1. Field of the Invention 5 The invention relates generally to the field of digital-to-analog conversion, and particularly to digital-to-analog conversion in radio transmitters. 2. Background Information Digital-to-Analog Converters (DACs) are often found in signal generation applications such as radio transmitters. See for example U.S. Patent No. 10 6,032,028 and U.S. Patent No. 5,010,585. As shown in FIGS. 1A-1D, the signal at the analog output of a DAC contains not only the desired signal, but also alias signals (or spectral copies) which have to be removed in a reconstruction filter, as shown for example in FIGS. 1A-1D. 15 As shown in FIG. 1A, a digital input signal is provided to an input 102 of the DAC 104. An output 106 from the DAC 104 provides an analog signal to an input of a reconstruction filter 108, and the reconstruction filter 108 outputs a reconstructed analog signal on its output 110. FIG. IB shows a magnitude spectrum of the digital signal provided to the input 102 of the DAC 104. 20 The spectral copies in the output from the DAC 104, fall off in amplitude as (sin(x)Vx due to the zero-order hold (ZOH) of the DAC. This can be seen in FIG. IC, which shows a magnitude spectrum of the signal output from the DAC 104. The sine response 142 behavior of the magnitude spectrum of the signal, which is due to the ZOH of the DAC, can be seen in FIG. IC. FiG. IC also -> <; shows an ideal frequency response 140 of the reconstruction filter 108. FIG. ID shows a magnitude spectrum of the signal output in the transmit frequency spectrum from the reconstruction filter 1 10, when the reconstruction filter 1 10 operates in accordance with the ideal frequency response 142. For performance reasons, the DAC outputs a desired signal at a fairly low frequency. Consequently, the reconstruction filter 108 is often implemented using discrete LC technology. If high filter performance is required, the filter will often be bulky and require trimming. In a typical radio transmitter known in the prior art, the output signal from the DAC is frequency upconverted in a mixer after passing through the reconstruction filter. An example of this is shown in FIG. 2, where an output of the reconstruction filter 108 is provided to a mixer 212. The mixer 212 mixes the output from the reconstruction filter 108 with a signal from a local oscillator 214. The frequency of the signal from the local oscillator 214 is represented as f L0 . The output of the mixer 212 is provided to an IF filter 216. In typical frequency mixers, the input signal is not only frequency- translated with the local oscillator frequency f L0 , but is also frequency-translated with its odd harmonics. Any of the spectral copies that survive the reconstruction filter 110, or that are present in circuit configurations where the signal from the DAC 104 is provided directly to the mixer 212 {i.e. , no reconstruction filtering is performed) can cause unacceptable distortion in or close to the signal band of interest. The distortion occurs because the spectral copies mix with the fundamental or harmonic frequencies of the local oscillator 214. FIGS. 3A, 3B show an example of this distortion. FIG. 3A shows a magnitude spectrum of a signal provided from the DAC 104 directly to the mixer 212 (where no reconstruction filtering is performed). FIG. 3B shows a magnitude spectrum of an output signal from the mixer 212, which results when the signal of FIG. 3A and the signal from the local oscillator 214 are input to the mixer 212. For the sake of illustration, the harmonics of the local oscillator 214 are assumed to have the same magnitude level as the fundamental frequency f L0 of the local oscillator 214. If the mixer 212 is well balanced, then onlv odd-order harmonics are of practical importance. These odd-order harmonics fall off in amplitude approximately following a Fourier series expansion of a square wave. The spectrum of the output of the mixer is simply found by displacing the double-sided spectrum of the output signal from the DAC, by all possible integer multiples (positive and negative) of the local oscillator frequency f L0 . For these reasons, the reconstruction filter 108 needs to be placed before the mixer 212. The filter requirements will become dependent on the frequency plan of the transmitter. In narrowband transmitters, its is possible through careful frequency planning to prevent frequency-converted spectral copies from appearing inside the frequency band of interest. For transmitters having bands wider than narrowband transmitters, effective frequency plans are very difficult to achieve due to the larger bandwidth of the signal. Moreover, high performance DACs usually have current source output, and suffer from decreasing linearity as the voltage swing over the load of the DAC increases. It is hence beneficial to keep the load impedance as low as possible, in order to help the DAC perform in as linear a fashion as possible. On the other hand, the linearity of some types of mixers is enhanced or benefitted by driving the mixer with a current source {i.e. , high impedance source). However, it is not possible to provide both a low impedance load for the DAC, and a high impedance source for the mixer, without signal attenuation which in turn degrades the attainable signal-to-noise ratio (SNR). In summary, the following problems exist in prior art systems: 1) a bulky and often expensive reconstruction filter is required in front of the mixer, and 2) the reconstruction filter cannot present both an optimal load for the DAC, and an optimal source for the mixer, without causing signal loss. Accordingly, a need exists for a system to overcome these problems. SUMMARY OF THE INVENTION In accordance with exemplary embodiments of the invention, a circuit is presented that eliminates the need for a bulky filter between the DAC and the mixer, and provides both a very low load impedance to the DAC, and a high source impedance to the mixer. In accordance with an exemplary embodiment of the invention, the DAC is connected directly to the mixer, and an IF filter after the frequency upconversion {i.e. , located downstream of the mixer) is used as a reconstruction filter. In addition to eliminating the bulky prior art reconstruction filter between the DAC and the mixer, a direct connection between the DAC and the mixer provides the DAC with a low-impedance load, and provides the mixer with a high-impedance source because the DAC and the mixer are effectively current-matched. In accordance with an embodiment of the invention, the update rate of the DAC, f DAC , is related to the frequency of the local oscillator by the following relation: f L0 = (n/2)(f DΛC ), where n is any positive integer. In an exemplary embodiment of the invention, n= l so that f L0 = (f DAC )/2. BRIEF DESCRIPTION OF THE DRAWINGS Other objects and advantages of the present invention will become apparent to those skilled in the art from the following detailed description of preferred embodiments, when read in conjunction with the accompanying drawings wherein like elements have been designated with like reference numerals and wherein: FIG. 1A shows a prior art circuit arrangement. FIGS. IB-ID show magnitude spectra of signals at various points of the FIG. 1 A circuit arrangement. FIG. 2 shows a prior art circuit arrangement. FIG. 3A shows a magnitude spectrum of an output signal from the DAC shown FIG. 2. FIG. 3B shows an exemplary magnitude spectrum of an output signal from the mixer shown in FIG. 2, when the reconstruction filter is omitted. FIG. 4A shows a circuit arrangement in accordance w ith an exemplary embodiment of the invention. FIG. 4B shows a circuit arrangement in accordance with an exemplary embodiment of the invention. FIG. 5A shows an exemplary magnitude spectrum of a signal present at an input of the mixer shown in FIGS. 4A, 4B. FIG. 5B shows an exemplary magnitude spectrum of a signal present at an output of the mixer shown in FIGS. 4A, 4B. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 4A illustrates an exemplary embodiment of the present invention, wherein a digital signal is provided to an input 402 of a DAC 404. An output 406 of the DAC 404 connects to an input of a mixer 412, and an output of the mixer 412 connects to an input of an intermediate frequency (IF) filter 418. The IF filter 418 outputs the filtered analog signal. In accordance with an exemplary embodiment of the invention, the IF filter 418 is used in place of a reconstruction filter. In accordance with an exemplary embodiment of the invention, the mixer 412 output frequency is used as an Intermediate Frequency, which is processed further (mixed again), and/or used as an output and later power amplified. A periodic signal 430 having a frequency f DAC is input to the DAC 404. The frequency f DAC reflects the update rate of the DAC 404. A periodic signal 432 having a frequency f L0 is provided to the mixer 412. The frequency f L0 is an integer multiple of (f DAC )/2. The periodic signal 432 having the frequency f L0 is provided to the mixer 412 for mixing with the analog signal output from the DAC 404. In an exemplary embodiment of the invention, the frequency f L0 is an odd- integer multiple of (f DAC )/2, for example an integer that is a member of the set {1 , 3, 5, 7, 9, 11 , ... }. In an exemplary embodiment of the invention, f L0 = (f DΛC )/2. The DAC 404 and the mixer 412 can be any appropriate DAC or mixer, and can be equivalent to or the same as DACs and mixers used in prior art circuit arrangements. The signal 432 having frequency f L0 that is provided to the mixer 412, can be a sin wave, a square wave, or other periodic signal waveform, that is acceptable to the mixer 412. In accordance with an exemplary embodiment of the invention, the waveform of the signal 432 depends on the type of the mixer 412. For example, passive mixers can require high level (sinusoidal) signals to switch the diode ring of the passive mixer, while active mixers can function with lower level signals that are sufficient to switch a Gilbert cell. The mixer 412 can be any type of mixer, including mixers other than multiplying-type mixers. The signal 430 having frequency f dac provided to the DAC 404, can be a sin wave, a square wave, or other periodic signal waveform, that forms an acceptable clock signal (update rate signal) for the DAC 404. In an exemplary embodiment of the invention, the signals 430, 432 having the frequencies f L0 and f DAC are square waves. In accordance with another exemplary embodiment, the waveform of the periodic signal 430 is different from the waveform of the periodic signal 432. FIG. 4B illustrates an exemplary embodiment of the present invention, wherein a local oscillator 414 is provided. The local oscillator 414 provides the update rate signal 430 to the DAC 404. Thus the frequency f DAC of the signal 430 output from the local oscillator 414 reflects the update rate of the DAC 404. A frequency changer 416 is also provided, to change the frequency (and optionally the waveform) of the signal 430 from the local oscillator 414, so that the signal 432 output by the frequency changer 416 has a frequency equal to (f DAC )/2. The signal 432 from the frequency changer 416 is provided to the mixer 412 for mixing with the analog signal output from the DAC 404. In an exemplary embodiment of the invention, the frequency changer 416 is not a tank circuit. The local oscillator 414 can be any appropriate oscillator providing an appropriate signal, and the frequency changer 416 can be any appropriate frequency changer or changer mechanism, including corresponding mechanisms known in the art. For example, the frequency changer 416 can be a commercially available frequency divider, or a commercially available frequency converter. With respect to both FIG. 4A and FIG. 4B, any appropriate subcircuit arrangement can be used, that will provide the mixer 412 with a periodic signal having a frequency that is an integer multiple of half the frequency of the periodic signal provided to the DAC 404. For example, two separate oscillators can be provided, each having an appropriate frequency output in accordance with the invention. In an embodiment of the invention, two such separate oscillators are synchronized so their output signals are in phase. In another embodiment, the two separate oscillators are not synchronized. The frequency changer 416 can be arranged so its output signal is either in phase, or out of phase, with its input signal. In an exemplary embodiment of the invention, the output signal of the frequency changer 416 is out of phase with its input signal. The signal 432 having frequency f L0 that is provided to the mixer 412, can be a sin wave, a square wave, or other periodic signal waveform. The signal 430 having frequency f Uat provided to the DAC 404, can be a sin wave, a square wave, or other periodic signal waveform. The frequency changer 416 can be configured to also change the waveform of the signal 430 that it receives from the local oscillator 414, for example from a square wave to a sin wave, or from a sin wave to a square wave. In accordance with the invention, under the condition that the highest frequency component in the desired signal (within the Nyquist band) is lower than (f L0 )/2 (which is the same as (I DAC ^ , no in-band distortion of the signal spectrum will occur. The reason is, that all spectral copies in the mixing process will either end up outside the signal band, or will superimpose on the signal itself. The spectral copies which do fall in the signal band will either add in-phase or in anti- phase to the signal. This phase shifting of the spectral bands is caused by the linear phase characteristics of the DAC's ZOH, which causes a phase difference of 180 degrees between spectral copies spaced f DAC apart. Mixing with different {e.g. , odd) harmonics of the local oscillator will not cause any phase difference because all harmonics are in phase with the fundamental, since the local oscillator can be modeled as a square wave. Since the phase relationship between the superimposed spectra are therefore constant and either a) zero degrees or b) 180 degrees, the only effect will be a minor change in signal amplitude. An example of the superposition of frequency converted spectral copies is shown in FIGS. 5A, 5B. In particular, FIG. 5A shows a magnitude spectrum of the signal at the input to the mixer 412. The magnitudes have a sine response, i.e., (l/x)sin(x), due to the ZOH of the DAC 404. FIG. 5B shows a magnitude spectrum of the signal at the output of the mixer 412. The segment blocks shown in FIG. 5A are pictorially stacked in FIG. 5B, to explicitly show where the segment blocks or signal spectra of FIG. 5A are relocated in FIG. 5B. Note that as shown in FIGS. 5A, 5B, the highest frequency component in the desired signal is exactly (f L0 )/2 = (f DAC )/4, which causes the out-of-band frequency-converted spectral copies to end up adjacent to the desired signal. In practice, the highest frequency component can be lower, so as to provide a guard band that allows the removal of the out-of-band distortion in the IF filter. The adjacent band, however, originates from mixing with the second harmonic and is therefore in practice of much lower amplitude than the desired signal. In summary, the present invention provides many advantages over prior art solutions. For example, a bulky reconstruction filter upstream of the mixer is eliminated, thus reducing cost and size of the resulting circuit. Frequency planning is simplified for both narrowband and wideband transmitters. One signal source can be used to generate both the DAC clock and the local oscillator signal that is provided to the mixer, thus reducing circuit complexity. When one signal is used to generate both the DAC clock and the local oscillator signal f L0 , a potential noise advantage can be achieved by canceling signal generator jitter in the IF filter output signal. In addition, linearity or linear behavior of the DAC is improved by providing a low impedance load for the DAC without causing signal loss. Linearity or linear behavior of the mixer is also improved by providing a high impedance source for the mixer without causing signal loss. It will be appreciated by those skilled in the art that the present invention can be embodied in other specific forms without departing from the spirit or essential characteristics thereof, and that the invention is not limited to the specific embodiments described herein. The presently disclosed embodiments are therefore considered in all respects to be illustrative and not restrictive. The scope of the invention is indicated by the appended claims rather than the foregoing description, and all changes that come within the meaning and range and equivalents thereof are intended to be embraced therein.

Claims: CJaims: 1. An apparatus comprising: a digital-to-analog converter; and a mixer for receiving an output of the digital-to-analog converter and for receiving a local oscillator signal, wherein a frequency of the local oscillator signal is equal to an integer multiple of half of an update rate of the digital-to-analog converter. 2. The apparatus of claim 1 , wherein the frequency of the local oscillator signal is equal to half of the update rate of the digital-to-analog converter. 3. The apparatus of claim 1 , further comprising: a signal generator which generates an output signal, wherein the signal generator output signal is used to generate the local oscillator output and the update rate of the digital-to-analog converter. 4. The apparatus of claim 1 , further comprising: an intermediate frequency filter for receiving an output of the mixer, wherein the intermediate frequency filter functions as a reconstruction filter. 5. The apparatus of claim 1, wherein the apparatus is in a wideband transmitter. 6. The apparatus of claim 1 , wherein the apparatus is in a narrowband transmitter. 7. The apparatus of claim 1 , wherein the mixer forms a low impedance load to the digital-to-analog converter, and the digital-to-analog converter forms a high impedance source to the mixer. 8. The apparatus of claim 1 , wherein the output of the digital-to-analog converter is connected directly to an input of the mixer. 9. The apparatus of claim 1 , wherein the frequency of the local oscillator signal is equal to an odd-integer multiple of half of the update rate of the digital-to- analog converter. 10. An apparatus comprising: a digital to analog converter; and a mixer for receiving an output of the digital to analog converter and for receiving a local oscillator signal, wherein the output of the digital-to-analog converter is connected directly to an input of the mixer. 11. The apparatus of claim 10, wherein a frequency of the local oscillator signal is equal to half of an update rate of the digital-to-analog converter. 12. The apparatus of claim 10, wherein a frequency of the local oscillator signal is equal to an odd-integer multiple of half of an update rate of the digital-to- analog converter 13. The apparatus of claim 10, further comprising: an intermediate frequency filter for receiving an output of the mixer, wherein the intermediate frequency filter functions as a reconstruction filter. 14. The apparatus of claim 10, wherein the apparatus is in a wideband transmitter. 15. The apparatus of claim 10, wherein the apparatus is in a narrowband transmitter. 16. The apparatus of claim 10, wherein a low impedance load is provided to the digital-to-analog converter and a high impedance source is provided to the mixer. 17. The apparatus of claim 16, wherein the high impedance source forms a low impedance load to the digital-to-analog converter, and the digital-to-analog converter forms a high impedance source to the mixer. 18. The apparatus of claim 10, further comprising: a signal generator which generates an output signal, wherein the signal generator output signal is used to generate the local oscillator signal and the update rate of the digital-to-analog converter. 19. A method for processing a digital signal, comprising: converting the digital signal to an analog signal based on the digital signal and a first periodic signal; and mixing the analog signal with a second periodic signal to form a second analog signal; wherein the second periodic signal has a frequency that is an integer multiple of half the frequency of the first periodic signal. 20. The method of claim 19, wherein the second periodic signal has a frequency that is an odd-integer multiple of half the frequency of the first periodic signal. 21. The method of claim 19, wherein the second periodic signal has a frequency that is half the frequency of the first periodic signal. 22. The method of claim 19, further comprising: filtering the second analog signal with an intermediate frequency filter. 23. The method of claim 19, wherein the step of converting is performed by a digital-to-analog converter whose update rate is determined by the first periodic signal.

Inventors: ERIKSSON PATRIK, PETERSSON PETER, ERIKSSON, PATRIK, PETERSSON, PETER
Applicants: ERICSSON TELEFON AB L M, TELEFONAKTIEBOLAGET LM ERICSSON (PUBL)
Application year: 2001
Main classification: H03M1
IPC codes: H03M1