ARRANGEMENTS AND METHODS FOR SPACE-TIME TURBO TRELLIS CODING

Abstract: First and second recursive space-time trellis coded modulation (STTCM) coders each produce, in successive symbol intervals, a plurality of T M-PSK symbols from a group of b input bits, where M=2<b>. The groups of b bits are supplied directly to the first coder and via an interleaver to the second coder, whose output symbols are conversely de-interleaved. A selector supplies T symbols from the first coder and T symbols from the de-interleaver alternately in successive symbol intervals to T output paths for T antennas providing transmit diversity. STTCM codes are described for different values of T and M and for fast and slow fading channels.

Description: ARRANGEMENTS AND METHODS FOR SPACE-TIME TURBO TRELLIS CODING This invention relates to coding for communications systems, for example for a cellular wireless communications system, for providing space-time (ST) diversity for so-called turbo trellis coding (TC) or trellis coded modulation (TCM) . Background of the Invention As is well known, wireless communications channels are subject to time-varying multipath fading, and it is relatively difficult to increase the quality, or decrease the effective error rate, of a multipath fading channel. One technique which has been found to be advantageous is antenna diversity, using two or more antennas (or signal polarizations) at a transmitter and/or at a receiver of the system. In a cellular wireless communications system, each base station typically serves many remote (fixed or mobile) units and its characteristics (e.g. size and location) are more conducive to antenna diversity, so that it is desirable to implement antenna diversity at least at a base station, with or without antenna diversity at remote units. At least for communications from the base station in this case, this results in transmit diversity, i.e. a signal is transmitted from two or more transmit antennas . S. M. Alamouti, "A Simple Transmit Diversity Technique for Wireless Communications", IEEE Journal on Selected Areas in Communications, Vol. 16, No. 8, pages 1451- 1458, October 1998 describes a simple transmit diversity scheme using space-time block coding (STBC) . For the case of two transmit antennas, complex symbols sO and -si* are successively transmitted from one antenna and simultaneously complex symbols si and sO* are successively transmitted from the other antenna, where * represents the complex conjugate. These transmitted symbols constitute what is referred to as a space-time block. It is also known to use various coding schemes in order to enhance communications. Among such schemes, it has been recognis.ed that so-called turbo coding (parallel concatenated convolutional coding) enables iterative decoding methods to achieve results which are close to the Shannon limit for A GN (additive white Gaussian noise) communication channels. A turbo coder uses two, typically identical, recursive systematic convolutional (RSC) component coders, signals to be transmitted being supplied directly to one of the component coders and via an interieaver to the other of the component coders . Accordingly, it would be desirable to combine turbo and space-time coding techniques in the same transmitter. V. Tarokh et al., "Space-Time Codes for High Data Rate Wireless Communication: Performance Criterion and Code Construction", IEEE Transactions on Information Theory, Vol. 44, No. 2, pages 744-765, March 1998 describes various convolutional, or trellis, codes which can be used with two or more transmit antennas to provide the advantages of trellis (convolutional) coding and space-time coding. Although these codes are considered optimal for maximum diversity gain, they are not necessarily optimal for coding gain. Furthermore, these codes are non-recursive. In contrast, it is well established that the best efficiency for turbo coding is achieved using recursive codes. Consequently, the codes described by Tarokh et al . are not particularly suitable for use in a- turbo coding arrangement. P. Robertson et al . , "Bandwidth-Efficient Turbo Trellis-Coded Modulation Using Punctured Component Codes", IEEE Journal on Selected Areas in Communications, Vol. 16, No. 2, pages 206-218, February 1998 describes a turbo coder using Ungerboeck and multidimensional TCM component codes, in which the interieaver operates on groups each of m information bits . For each step corresponding to a group of m information bits, a signal mapper associated with each component coder produces n symbols, where n=D/2 and D is the signal set dimensionality; for example D=2 or 4 and n=l or 2. An n-symbol de-interleaver de-interleaves output symbols from the second component coder, and a selector alternately for successive steps selects symbols output from the first component coder and symbols from the de- interieaver and supplies them to a single output path. This arrangement does not provide transmit diversity and this document is not concerned with space-time coding. G. Bauch, "Concatenation of Space-Time Block Codes and "Turbo"-TCM" , Proceedings of the International Conference on Communications, ICC 99, pages 1202-1206, June 1999 describes two types of turbo trellis coded modulation (TCM) coder, whose output is supplied to a space-time block coder, so that the turbo-TCM and STBC arrangements are simply concatenated with one another. One of these two types of turbo TCM coder is as described by Robertson et al . (to which reference is made for details) as discussed above using Ungerboeck codes and providing one symbol at the output of the mapping function, but the Bauch illustration of this does not show the symbol de- interleaver. This Bauch publication does not discuss multidimensional component codes. A continuing need exists to provide further improvements through coding in wireless communications. Summary of the Invention According to one aspect, this invention provides a coding arrangement comprising: first and second recursive STTCM (space-time trellis coded modulation) coders each arranged to produce, in each of a plurality of successive symbol intervals, a plurality of T M-PSK (M-ary phase shift keying, where M is a plural integer) symbols from b bits supplied thereto, where b is an integer; an interieaver arranged to interleave groups each of b input bits within an interleaving block with a mapping of even-to-even and odd-to- ' odd, or even-to-odd and odd-to-even, positions; input bits supplied to the first coder and to the interieaver, and interleaved bits supplied from the interieaver to the second coder; a symbol de-interleaver arranged to de-interleave, in a manner converse to the interleaving by the interieaver, groups of T symbols produced by the second STTCM coder; and a selector arranged to supply T symbols produced by the first STTCM coder and T symbols from the de-interleaver in alternating symbol intervals to respective ones of T output paths . Preferably each coder is arranged to produce, in each • symbol interval, T modulo-M sums of linear combinations of current and one or more preceding groups of the b bits supplied to the coder to constitute the T symbols produced by the coder in the respective symbol interval. Conveniently, for each coder, M=2 b and b is at least 2; for example b=2 or 3. The T output paths typically lead to T antennas of a transmitter thereby providing space-time diversity for turbo coded TCM symbols, and for example T=2 , 3, or . The invention also provides a method of coding for providing space-time diversity for information to be transmitted from a plurality T of antennas, comprising the steps of: in each of a plurality of successive symbol intervals, producing T symbols at outputs of each of first and second recursive STTCM (space-time trellis coded modulation) coders, to the first of which coders input bits are supplied directly and to the second of which coders said information bits are supplied after interleaving of bit groups for respective symbol intervals in an interleaving block with a mapping of even-to-even and odd-to-odd, or even-to-odd and odd- to-even, positions; de-interleaving, in a manner converse to the interleaving, groups of T symbols produced by the second STTCM coder; and selecting the T symbols produced by the first coder and the T symbols from the de-interleaving step in respective first and second alternating symbol intervals for supply to paths to the T antennas . The preferred features may be combined as appropriate, as would be apparent to a skilled person, and may be combined with any of the aspects of the invention. Brief Description of the Drawings The invention will be further understood from the following description with reference to the accompanying drawings, in which by way of example: Fig. 1 illustrates parts of a known space-time block code (STBC) transmitter; Fig. 2 illustrates a known turbo coder; Fig. 3 illustrates parts of a turbo space-time trellis coded modulation (STTCM) coding arrangement for a transmitter using two transmit antennas, in accordance with an embodiment of this invention; Fig. 4 illustrates a 16-state recursive trellis coder which can be used in the arrangement of Fig. 3; and Fig. 5 illustrates a decoding arrangement for use with the coding arrangement of Fig. 3. Detailed Description Referring to the drawings; Fig. 1 illustrates parts of a known space-time block code (STBC) transmitter. For simplicity and clarity in this and other figures of the drawings, only those parts are shown which are necessary for a full understanding of the prior art and embodiments of this invention. The transmitter of Fig. 1 includes a serial-to- parallel (S-P) converter 10, an M-PSK mapping function 12, and a space-time block coder (STBC) 14 providing outputs, via transmitter functions such as up-converters and power amplifiers not shown but represented in Fig. 1 by dashed lines, to at least two antennas 16 and 18 which provide transmit diversity. The S-P converter 10 is supplied with input bits of information to be communicated and produces output bits on two or more parallel lines to the M-PSK mapping function 12, which produces from the parallel bits sequential symbols i, x 2 , ... of an equal-energy signal constellation. For example, as shown in Fig. 1 the mapping function 12 may provide a Gray code mapping of in each case 2 input bits from the S-P converter 10 to respective ones of M=4 signal points of a QPSK (quadrature phase shift keying) signal point constellation. It can be appreciated that the mapping function 12 can provide any desired mapping to a signal point constellation with any desired number M of phase states; for example M=2 (for which the S-P converter 10 is not required) , 4, or 8. The QPSK symbols i, x 2 , ... , represented by complex numbers, are supplied to the STBC 14, which for simplicity is shown in Fig. 1 as having two outputs for the respective transmit antennas 16 and 18, but may instead have more than two outputs for a corresponding larger number of transmit antennas . For the case of two antennas as shown, the STBC 14 forms a space-time block of symbols, as represented in Fig. 1, from each successive pair of symbols Xi and x supplied to its input. More particularly, the STBC function is represented by a T-by-T orthogonal matrix H x , where T is the number of transmit antennas and hence symbol outputs of the STBC 14. For the case of T=2 as represented in Fig. 1, X-, H X ( X 1 'X 2 )= -x, In accordance with this matrix H x , for each pair of PSK symbols Xi and x 2 supplied to the input of the STBC 14, in a first symbol interval the antenna 16 is supplied with the symbol Xi and the second antenna 18 is supplied with the symbol x 2/ and in a second symbol interval the first antenna 16 is supplied with the symbol -x 2 * and the second antenna 18 is supplied with the symbol Xi*, where * denotes the complex conjugate. Thus both QPSK symbols in each pair are transmitted twice in different forms, from different antennas and at different times to provide both space and time diversity. It can be seen that each column of the matrix H x indicates the symbols transmitted in successive intervals from a respective antenna, and each row represents a respective symbol transmission interval. Referring to Fig. 2, a known turbo (parallel concatenated convolutional) coder comprises two recursive systematic convolutional (RSC) coders 20 and 22 which are referred to as the constituent or component coders of the turbo coder, an interieaver 24, and a selector 26. Input bits are supplied to the input of one coder 20, which produces at its outputs both systematic bits SI, which are the same as the input bits, and parity bits Pi. The input bits are also supplied to and interleaved by the interieaver 24, and the interleaved bits are supplied to the input of the other coder 22, which produces at its outputs both systematic bits S2, which are the same as the interleaved input bits, and parity bits P2. The outputs of the two coders 20 and 22 are supplied to inputs of the selector 26, except that typically and as shown in Fig. 3 the systematic bit output of the coder 22 is not connected because the interleaved bits at this output are never selected by the selector 26. The selector 26 selects all of the systematic bits SI, and some or all of the parity bits Pi and P2 from the coders 20 and 22 respectively, and supplies them to an output of the turbo coder as output bits. The selection of parity bits depends upon the rate of the coder. For example, for a rate 1/3 (3 output bits for each input bit) coder, the selector 26 can select all of the parity bits PI and P2. For a rate 1/2 (2 output bits for each input bit) coder, the selector 26 can alternately select the parity bits PI and P2 , so that only half of the parity bits PI and half of the parity bits P2 are output, this process being referred to as puncturing. In the turbo-TCM arrangement (Robertson et al . ) referred to in the Background of the Invention, the interieaver 24 operates on groups each of m bits which are mapped at the output of each component coder into a PSK symbol combining the systematic and parity information. The symbols from the second component coder are de-interleaved by a symbol de-interleaver, and the output selector alternately selects the symbols output from the first component coder (and the de-interleaver. The interieaver (and consequently also the de-interleaver) in this case must provide an even-to-even and odd-to-odd (or even-to- odd and odd-to-even) position mapping. In the concatenated SBTC and turbo code (Bauch) arrangements referred to in the Background of the Invention, in essence the output bits of a turbo coder such as that of Fig. 3 are supplied as input bits to a space-time block coder such as that of Fig. 1, or the output symbols from a turbo-TCM coder as described by Robertson et al. are supplied as input symbols to an STBC coder 14 as described above with reference to Fig. 1. Fig. 3 illustrates parts of a turbo space-time trellis coded modulation (STTCM) coding arrangement for a transmitter using two transmit antennas, in accordance with an embodiment of this invention. As in the case of Fig. 1, the two antennas are referenced 16 and 18, and input bits of information to be communicated are supplied to the S-P converter 10, which has b outputs for groups each of b information bits. The remainder of Fig. 3 represents the turbo STTCM coding arrangement, which comprises first and second recursive STTCM component coders 30 and 32, an interieaver 34, a symbol de-interleaver 36, and a selector 38 having two outputs for the respective transmit paths to the two antennas 16 and 18. The coders 30 and 32 and the interieaver 34 each have b inputs for the groups of information bits . The groups of b bits supplied from the S-P converter 10 are interleaved in these groups by the interieaver 34. The non-interleaved bit groups supplied to the coder 30, and the interleaved bit groups supplied to the coder 32, are coded and mapped into symbols by these functions as described further below. For the case of two output paths corresponding to the two transmit antennas as shown in Fig. 3, each of the coders 30 and 32 produces two M-PSK symbols in respect of each bit group, where M=2 b . The symbols produced in each symbol interval by the first coder 30 from the non-interleaved bit groups are identified as xlχ and xl 2 as shown in Fig. 3. The symbols produced by the second coder- 32 are de-interleaved by the de- interleaver 36, which operates conversely to the interieaver 34, to produce symbols in each symbol interval which are identified as x2ι and x2 2 as shown in Fig. 3. It is assumed here for convenience and simplicity that the coders 30 and 32 are identical, but as for known turbo coders this need not necessarily be the case and the coders 30 and 32 could instead differ from one another. The selector 38 is controlled by a control signal of alternating ones and zeros (1010... as illustrated) at the bit group rate, and performs selection and puncturing functions as represented in Fig. 3 by switches within the selector 38. In a first state of the control signal, for example when the control signal is a binary 1, the switches of the selector 38 have the states illustrated in Fig. 3 in which the symbols xli and xl 2 from the coder 30 are supplied as symbols x 1 and x 2 respectively to the output paths to the transmit antennas 16 and 18 respectively, and the symbols x2χ and x2 2 from the symbol de- interleaver 36 are not used. In a second state of the control signal, for example when the control signal is a binary 0, the switches of the selector 38 have their opposite states in which the symbols x2ι and x2 from the symbol de-interleaver 36 are supplied as the symbols x 1 and x 2 respectively to the output paths to the transmit antennas 16 and 18 respectively, and the symbols xlχ and xl 2 from the coder 30 are not used. It can be appreciated that, with the selector 38 alternately selecting non-interleaved symbols from the coder 30 and de-interleaved symbols from the de-interleaver 36 as described above, it is necessary (for decoding reasons as explained in the Robertson et al . publication referred to above) for the interieaver 34 to map even positions at its input to even positions at its output, and odd positions at its input to odd positions at its output (or, alternatively, even- to-odd and odd-to-even position mapping) , as in the case of the Robertson et al . arrangement discussed above. The interieaver 34 is arranged to provide such mapping accordingly, and the de- interleaver 36 provides a converse mapping as described above. The coding arrangement of Fig. 3 is further described below in terms of a simple example, in which the component coders -30 and 32 are assumed to be identical 8-state recursive QPSK (i.e. M=4) STTCM coders, with two output paths for two transmit antennas, and an interleaving block size of 6 groups each of b=2 bits. The coder states are numbered 0 to 7, and each QPSK symbol has one of four states numbered 0 to 3. Each of the coders 30 and 32 operates in accordance with the following Table 1, in which the next state of the coder and the two output QPSK symbols are dependent upon the current state of the coder and the b=2 input bits of the current grou . These are represented in the table in the form p/qr, where p denotes the next state of the coder and q and r represent the two output QPSK symbols, e.g. the symbols li and xl 2 respectively for the coder 30: Table 1 For example, if an input bit sequence to this coding arrangement is c= (10, 00, 11, 10, 01, 01) , then from Table 1 it can be seen that, starting from state 0, the coder 30 generates the symbol sequence (xl 1 xl 2 ) ={ (0,2) , (1, 1) , (1,3) , (2, 1) , (0, 1) , (0, 0) } with its next states being successively 4, 6, 7, 7, 2, and 5. If the interieaver 34 interleaves the sequence c to form the interleaved sequence cι= (01, 10 , 10, 01, 11, 00) , i.e. if the interieaver moves 2-bit groups in positions numbered 0 to 5 in the interleaving block to positions 2, 5, 4, 1, 0, and 3 respectively, then the coder 32 generates the symbol sequence {(2,2), (3,2), (0,3), (3,0), (3,1), (1,2)} from an initial state of 0 with its next states being successively 1, 5, 3, 4, 3, and 5. The de-interleaver 34 de-interleaves this sequence to produce the sequence (x2ι,x2 2 ) ={ (0, 3 ) , (1, 2 ) , (3 , 1) , (3 , 2) , (2 , 2) , (3 , 0) } . Consequently, the selector 38 produces the symbol sequence x 1 = (0, 1, 1, 3, 0, 3) on the output path to the antenna 16 and the symbol sequence x 2 = (2, 2 , 3, 2 , 1, 0) on the output path to the antenna 18. From the above description and from Fig. 3 it can be appreciated that the units 30, 32, 34, 36, and 38 provide a turbo coding arrangement for space-time trellis coded modulation, thereby enabling advantages of coding gain and diversity gain of these coding functions to be combined. While the above simple example serves to assist in providing a full understanding of the coding arrangement of Fig. 3 and its operation, it can be appreciated that selection of the particular STTCM codes to be used can have a significant affect on the performance of the coding arrangement. For example, the STTCM codes known from the Tarokh et al . publication referred to above are not recursive as is important to obtain the full advantages of a turbo coding arrangement, and are not necessarily optimal for coding gain. STTCM codes can be found through systematic code searching techniques (where this is computationally feasible) to provide a theoretically maximal diversity gain and an improved coding gain, as described by S. Baro et al . in "Improved Codes for Space-Time Trellis Coded Modulation", IEEE Communications Letters, Vol. 4, No. 1, pages 20-22, January 2000. Desirable STTCM codes (and consequently forms of the STTCM coders 30 and 32) can be determined in other ways either known or yet to be devised. The particular codes described below are given by way of example of recursive STTCM codes which are considered to provide advantageous performance in particular situations. Fig. 4 illustrates a 16-state QPSK recursive feedback STTCM coder which can be used to constitute each of the component coders 30 and 32 in the coding arrangement described above with reference to Fig. 3. Referring to Fig. 4, the recursive feedback STTCM coder comprises four delay elements 41 to 44 each providing a delay T corresponding to each group of b=2 (for QPSK) input bits c° and to the coder at a time t, the delay elements thereby providing the coder with 2 4 =16 states. In addition, the coder comprises adding elements 45 and 46, multiplication functions 47 to 52, and a summing function 53 which is supplied with the outputs of the multiplication functions 47 to 52 and which produces two output symbols (for two transmit antennas) x^ and x^ at the time t, corresponding for example to the output symbols li and xl 2 respectively of the coder 30 as described above . The input bit c° is supplied to one input, and the outputs of the delay elements 41 and 42 are supplied to other inputs, of the adding element 45, whose output is supplied to the input of the delay element 41. The output of the delay element 41 is also supplied to the input of the delay element 42. The outputs of the adding element 45 and of the delay elements 41 and 42 are also supplied to inputs of the multiplication functions 47 to 49 respectively, which are also supplied with multiplication coefficients ( aj , a 2 0 ) , ( , a x 2 ) , and ( a , ) respectively. Similarly, the input bit c^ is supplied to one input, and the outputs of the delay elements 43 and 44 are supplied to other inputs, of the adding element 46, whose output is supplied to the input of the delay element 43. The output of the delay element 43 is also supplied to the input of the delay element 44. The outputs of the adding element 46 and of the delay elements 43 and 44 are also supplied to inputs of the multiplication functions 50 to 52 respectively, which are also supplied with multiplication coefficients ( bj , ~ b 2 0 ) , ( b^ , b^ ) , and ( ^b j ) respectively. The coder of Fig. 4 provides a modulo-4 sum of the linear combinations of the current and delayed binary inputs , represented algebraically by the Equation: = ∑S° t - J0 „ + ∑ t-Λ mod 4 (1) where ne{l,2} identifies the two output symbols, the memory order v of the coder is given by v =v 0 +v 1 , a" . , b" e {0,1,2,3} for QPSK, j L e {θ,l, ..., Vi}, and a variable c is defined as fi = c t + ∑ fi j ι mod 2 with ie{θ,l}. The performance of STTCM coding on fast fading channels (channels for which the fading coefficient changes from one symbol interval to the next) is determined by the minimum symbol Hamming distance δe m i an< 3- the minimum product distance pd min . Table 2 lists recursive QPSK STTCM codes, in terms of their multiplication coefficients for various values v of memory order, which have been found to best satisfy these criteria and accordingly are presently preferred, using the structure exemplified by Fig. 4 as described above (contracted or expanded if appropriate for the respective memory order) : Table 2 Similarly, an 8-PSK STTCM coder for a transmitter with two transmit antennas and for fast fading channels - provides a modulo-8 sum of the linear combinations of the current and delayed binary inputs, c° , c , and at a time t, can be represented algebraically by the Equation: x, + Σc 2 .d n mod 8 (2) where again ne{l,2} identifies the two output symbols, the memory order v of the coder is given by v =v 0 +v 1 +v 2 , for 8-PSK a^b^d^ e{θ,l,2,...,7}, j ± ε {θ , 1, ..., v , and a variable cj: is Vl • defined as c^cJ + Vc^, mod 2 with ie{θ,l,2}. Table 3, having a similar form to Table 2 above, lists multiplication coefficients for presently preferred 8-PSK STTCM codes: Table 3 For slow fading channels (channels for which the fading coefficient is constant over the symbols of an interleaving block) , recursive feedback STTCM codes can be derived from feedforward codes by rearranging the order of outputs of the trellis. The following Table 4 represents, for the feedforward code and the derived recursive feedback code and in a similar manner to that of Table 1 above, the next state of the coder and the two output symbols of a 4-state QPSK STTCM coder for a coding arrangement for two transmit antennas: Table 4 Similarly, the following Table 5 represents, for the feedforward code and the derived recursive feedback code, the next state of the coder and the two output symbols of an 8-state QPSK STTCM coder for a coding arrangement for two transmit antennas : Table 5 Similarly, the following Table 6 represents, for the derived recursive feedback code only, the next state of the coder and the two output symbols of an 8-state 8-PSK STTCM coder for a coding arrangement for two transmit antennas : Table 6 The recursive feedback STTCM codes defined in Tables 4, 5, and 6 can be used with advantage for a transmitter with two transmit antennas for slow fading channels. The above examples relate to recursive STTCM codes that provide two output symbols for each group of b input bits, for supply to two antennas in an alternating manner from the two component coders 30 and 32 of the turbo coding arrangement of Fig. 3. It can be appreciated that the same principles can be applied to a coding arrangement for use with a greater number of transmit antennas by using recursive STTCM component codes that provide a correspondingly greater number of output symbols for each group of b input bits. Considered generally, if the transmitter has a plurality of T transmit antennas, then each of the two component coders is chosen to be a recursive STTCM coder providing T output symbols for each group of b input bits, and the selector alternately selects the T symbols of the two component coders for supply to T output paths for the T transmit antennas . By way of example, the following Table 7 lists, in a similar manner to Table 2 above, multiplication coefficients for various values v of memory order for recursive QPSK STTCM codes for three output symbols, i.e. for three transmit antennas: Table 7 For QPSK symbols for three transmit antennas for fast fading channels, the recursive STTCM coders are represented by Equation (1) given above but with ne {1,2,3} corresponding to the three output paths . By way of further example, the following Table 8 lists, in a similar manner, multiplication coefficients for various values v of memory order for recursive 8-PSK STTCM codes for four output symbols, i.e. for four transmit antennas: Table 8 For 8-PSK symbols for four transmit antennas for fast fading channels, the recursive STTCM coders are represented by Equation (2) given above but with ne{l,2,3,4} corresponding to the four output paths. The form of STTCM coders for other combinations of M-PSK symbols and numbers of transmit antennas can be seen from these examples . Fig. 5 illustrates a decoding arrangement for use in a receiver for receiving signals from a transmitter using a coding arrangement as described above with reference to Fig. 3. The receiver (not shown) may have a single receive antenna and related circuits, such as down converters and signal amplifiers and samplers, to provide received symbols which are supplied to the input of the decoding arrangement as described below, or it may have two or more receive antennas the signals from which are supplied to a correspondingly modified decoding arrangement . The decoding arrangement comprises a de-puncturing selector 60, two soft output trellis code decoders 61 and 62, symbol-based interleavers 63 and 64, and symbol-based de- interleavers 65 and 66. The de-interleavers 65 and 66 operate with the same symbol-based de-interleaving as the de- interleaver 36 of the turbo coding arrangement of Fig. 3, conversely to the interleaving operation of the interleavers 63 and 64 and, equivalently, the bit group interieaver 34 of the turbo coding arrangement . The decoding arrangement of Fig. 5 is a symbol-by- symbol log-MAP (maximum a posteriori) decoder, with respect to which reference is directed to the Robertson et al. publication referred to above, which contains a detailed discussion of such decoders. As is known, the decoders 61 and 62, which are complementary to the component coders 30 and 32 respectively of the turbo coding arrangement of Fig. 3, operate iteratively to take advantage of the turbo coding gain. Thus the decoder 61 operates on non-interleaved or de-interleaved information, and the decoder 62 operates on interleaved information, the interleavers 63 and 64 and the de-interleaver 65 providing the symbol-based interleaving and de-interleaving of information coupled to and between these decoders. After a desired number of iterations, an output decision is derived from the decoder 62 via the de-interleaver 66. Accordingly, in alternating symbol intervals determined by a control signal of alternating ones and zeros (1010...) supplied to the selector 60, received symbols r are supplied to the decoder 61 and are interleaved by the interieaver 63 to produce interleaved received symbols f which are supplied to the decoder 62. In a first iteration of the decoding arrangement, the decoder 61 determines extrinsic and systematic information (as described in the Robertson et al . publication, these are inseparable in a symbol-by-symbol decoding arrangement) Λι, es , constituting log-likelihood ratios for respective received symbols, which are interleaved by the interieaver 64 to produce interleaved extrinsic and systematic information Λ 1/es . This interleaved extrinsic and systematic information is supplied to the decoder 62, which uses it as a priori information to decode the interleaved received symbols r . The decoder 62 consequently produces interleaved extrinsic and systematic information Λ 2 , es a *id interleaved information Λ 2 representing the received symbols. The interleaved extrinsic and systematic information Λ 2 . es is de-interleaved by the de- interleaver 65 to produce extrinsic and systematic information Λ 2 , es which is supplied to the decoder 61 for use as a priori information in a second iteration of the decoding arrangement. This process is repeated for the desired number of iterations, after which the information Λ 2 produced by the decoder 62 is de-interleaved by the de-interleaver 66 to provide an output decision for the respective symbols. It is observed that, as described in the Robertson et al . publication, the decoders 61 ' and 62 are arranged to avoid using the same systematic information more than once in each iteration. Although particular embodiments of the invention are described in detail above, it can be appreciated that numerous modifications, variations, and adaptations may be made within the scope of the invention as defined in the claims . In particular, it is observed that other recursive STTCM component codes, bit group sizes, M-PSK symbols, and numbers of output paths or transmit antennas can be used.

Claims: WHAT IS CLAIMED IS: 1. A coding arrangement comprising: first and second recursive STTCM (space-time trellis coded modulation) coders (30, 32) each arranged to produce, in each of a plurality of successive symbol intervals, a plurality of T M-PSK (M-ary phase shift keying, where M is a plural integer) symbols from b bits supplied thereto, where b is an integer; an interieaver (34) arranged to interleave groups each of b input bits within an interleaving block with a mapping of even-to-even and odd-to-odd, or even-to-odd and odd- to-even, positions; input bits supplied to the first coder and to the interieaver, and interleaved bits supplied from the interieaver to the second coder; a symbol de-interleaver (36) arranged to de- interleave, in a manner converse to the interleaving by the interieaver, groups of T symbols produced by the second STTCM coder; and a selector (38) arranged to supply T symbols produced by the first STTCM coder and T symbols from the de-interleaver in alternating symbol intervals to respective ones of T output paths . 2. A coding arrangement as claimed in claim 1 wherein each coder is arranged to produce, in each symbol interval, T modulo-M sums of linear combinations of current and one or more preceding groups of the b bits supplied to the coder to constitute the T symbols produced by the coder in the respective symbol interval. 3. A coding arrangement as claimed in any one of claims 1-2 wherein, for each coder, M=2 b and b is at least 2. 4. A coding arrangement as claimed in claim 3 wherein b=2 or b=3. 5. A coding arrangement as claimed in any one of claims 1-4 wherein T=2 or T=3 or T=4. 6. A method of coding for providing space-time diversity for information to be transmitted from a plurality T of antennas, comprising the steps of: in each of a plurality of successive symbol intervals, producing T symbols at outputs of each of first and second recursive STTCM (space-time trellis coded modulation) coders (30, 32), to the first of which coders (30) input bits are supplied directly and to the second of which coders (32) said information bits are supplied after interleaving of bit groups for respective symbol intervals in an interleaving block (34) with a mapping of even-to-even and odd-to-odd, or even-to- odd and odd-to-even, positions; de-interleaving (36), in a manner converse to the interleaving, groups of T symbols produced by the second STTCM coder; and selecting (38) the T symbols produced by the first coder and the T symbols from the de-interleaving step in respective first and second alternating symbol intervals for supply to paths to the T antennas . 7. A method as claimed in claim 6 wherein in each symbol interval each recursive STTCM coder produces said T symbols as M-PSK (M-ary phase shift keying) symbols constituted by modulo- M sums of linear combinations of current and one or more preceding groups each of b bits supplied to the coder in the respective symbol intervals . 8. A method as claimed in any one of claims 6-7 wherein, for each coder, M=2 b and b is at least 2. 9. A method as claimed in claim 8 wherein b=2 or b=3. 10. A method as claimed in any one of claims 6-8 wherein T=2 or T=3 or T=4. 11. A method as claimed in any one of claims 6-10 wherein in each symbol interval each recursive STTCM coder produces said T symbols as M-PSK (M-ary phase shift keying) symbols from a current and one or more preceding groups each of b bits supplied to the coder in the respective symbol intervals, where M=2 b .

Inventors: CHEN ZHUO, FIRMANTO WELLY, VUCETIC BRANKA, YUAN JINHONG, CHEN, ZHUO, FIRMANTO, WELLY, VUCETIC, BRANKA, YUAN, JINHONG
Applicants: NORTEL NETWORKS LTD, NORTEL NETWORKS UK LTD, NORTEL NETWORKS LIMITED, NORTEL NETWORKS UK LIMITED
Application year: 2001
Main classification: H04L1
IPC codes: H03M13, H04L1, H04L27