HIGH POWER ELECTRONIC BALLAST WITH AN INTEGRATED MAGNETIC COMPONENT

Abstract: A high power electronic ballast for operating one or more discharge lamps includes a power factor correction high frequency SEPIC converter input stage in cascade with an output stage made up of a half-bridge resonant converter. In place of the conventional isolation transformer in the output stage, the input stage utilizes a single integrated transformer inductor magnetic component, thereby eliminating the usual output stage isolation transformer. The input stage is operated in a critical conduction mode which further improves the electronic ballast.

Description: High power electronic ballast with an integrated magnetic component The present invention relates to an electronic ballast apparatus for operation of gas discharge lamps, and more particularly to a high power electronic ballast equipped with a unique and improved circuit topology that provides electric isolation from the AC power line voltage. A conventional electronic ballast operated from a 60 Hz AC power line usually includes a two-stage converter circuit in order to supply a high frequency AC current to a load consisting of one or more gas discharge lamps In order to provide protection against the hazard of an electrical shock, the output stage of the converter circuit is coupled to the lamp load via an isolation transformer, thereby providing electπc isolation between the ballast output terminals and the converter circuit One example of an improved pπor art electronic ballast is descπbed in U S Patent 5,084,653, which issued 1/28/92 This electronic ballast provides electπc isolation in the front end stage of the apparatus This patent descπbes an electronic ballast for poweπng three seπes connected fluorescent lamps with a 30 kHz lamp current and compπsing a half- bπdge seπes-resonance type inverter circuit powered from a substantially constant magnitude DC supply voltage deπved from ordinary 60 Hz power line voltage by way of a bπdge rectifier and a single transistor DC-to-DC convertei using an energy stoπng inductor with an isolated secondary winding from which the DC supply voltage is deπved Thus, the DC supply voltage, the inverter circuit and the ballast output terminals are all electπcally isolated from the AC power line and thereby provide protection against the hazard of an electπc shock. The front end stage of this electronic ballast is based upon a flyback DC-to- DC converter This type of converter requires an energy-stoπng inductor which is a rather large and relatively expensive magnetic component, especially as compared with the inductors used in a boost converter or a single-ended primary inductor converter (SEPIC convertei) with the same or comparable power ratings Anothei disadvantage of the flyback converter is the large magnitude of its pulsating input current which usually requires larger EMI filters in order to achieve comparable EMC performance to that of a boost converter or a SEPIC converter Another prior at high frequency electronic ballast for gas discharge lamps is shown in the accompanying Fig. 1 of the drawings. This electronic ballast circuit basically consists of two building block:;. The front end is a boost converter for power factor correction and universal line voltage regulation. The main components are a transistor power switch Ql, an inductor LI, a diode D5 and the DC storage capacitor Cl along with an EMI filter and the diode bridge rectifier interposed between the AC supply voltage and the boost converter. The transistor switch Ql is periodically switched on and off by a control circuit, for example, a Motorola Corporation product MC34262, as a function of the voltage across capacitor Cl and the current flowing the transistor switch Ql. The back end is a typical voltage-fed half-bridge inverter loaded with a group of lamps via a resonant tank L2-C3. The main components are the power switches Q2 and Q3, resonant components including capacitor C3, inductor L2 and possibly the magnetizing inductance of the output transformer Tl. The capacitors Clp in the secondary circuit of the transformer Tl are usually provided in order to ballast the lamp current and to protect against possible lamp rectification at the end of lamp life. There are four magnetic components in the circuit configuration shown in Fig. 1, i.e. the EMI filter L0, the boost choke LI, the resonant inductor L2 and the output isolation transformer Tl. The operation of the power switches Q2 and Q3 is controlled by a high voltage control IC, for example, an IC UBA 2010 manufactured by Philips, as a function of current flow in transistor switch Q3 and the voltage on capacitor C3. Here too the size and cost of the magnetic components of the apparatus make this type of high frequency electronic ballast less than an optimal choice, especially from a competitive commercial viewpoint. It is therefore an object of the invention to provide a high power electronic ballast having a reduced number of magnetic components with a concomitant reduction in the size and cost of the electronic ballast. Another object of the invention is to provide a compact and inexpensive high power electronic ballast which uses a single integrated transformer-inductor combination in the ballast input stage that exhibits reduced power losses in the magnetic components. A further object of the invention is a new and improved method of operating a high power electronic ballast. More specifically, the coupled inductor based SEPIC converter is designed to be operated in a boundary or critical conduction mode with variable frequency during the 60Hz line cycle. The variable frequency range cound be, for example, from 50kHz to 150kHz. The control circuit along with the sensing arrangement of the invention leads to close to unity power factor at the line input terminal. The above and other objects and advantages are achieved in accordance with the present invention wherein the isolation transformer is introduced into the power factor correction front end stage of the electronic ballast as a single integrated magnetic component that provides the electπc isolation function as well as the converter inductor function. As a result, the output isolation transformer common in pnor art high frequency electronic ballast circuits is no longer required. The invention herein is based upon the following observations. In a pnor art electronic ballast of the type shown in Fig. 1, a major function of the output transformer Tl is to provide electπc isolation between the pπmary and the secondary in order to meet the ANSI safety requirements In current-fed half-bπdge converters, such an output transformer also serves as a resonant component and voltage gain booster But, in a voltage-fed half- bπdge inverter as shown in Fig. 1, the voltage gain can be obtained from the resonant tank consisting of inductor L2 and capacitor C3. Therefore, it is possible to remove the output transformer if the electπc isolation is introduced into the front end part of the circuit. This results in a single magnetic component AC-DC converter with a power factor correction (PFC) front end stage providing galvanic isolation in cascade with a single magnetic component half-bπdge resonant converter in accoi dance with the present invention. The introduction of an isolated power factor correction front end stage employing a single magnetic component eliminates the requirement for the output transformer. As a result, the total number of magnetic components of the ballast circuit is reduced to three and the overall size and weight of the magnetic components are less than those of pnor art electronic ballast circuits of the type descπbed above The related circuit losses and cost are reduced and further circuit miniatuπzation is now feasible In a preferred form of the invention, the front end stage of the high power electronic ballast utilizes a coupled inductor isolated SEPIC converter The invention also contemplates the use of an isolated boost circuit as the front end stage of the electronic ballast In another advantageous form of the invention we provide an improved control function for the transistor switch in the PFC input stage of the apparatus, 1 e the apparatus is operated in a so-called cπtical conduction mode in which the transistor switch is turned on as soon as the input inductor current falls to zero Pnor art converters usually operate in a continuous conduction mode or in a discontinuous conduction mode In a continuous conduction mode (CCM) apparatus, the circuit efficiency is generally high (low losses) but the control circuit is more complex and therefore costly. In the discontinuous conduction mode (DCM) a simple control circuit is possible, but the circuit losses are higher. The cntical conduction mode of the present invention has the advantage of a better tradeoff between the control complexity and the circuit losses than is possible with either of the two pnor art control methods, i.e CCM or DCM. The foregoing and other aspects of the invention will be descnbed and elucidated in greater detail with reference to the accompanying drawings, in which: Fig 1 shows a general circuit structure for a pnor art high frequency electronic ballast which utilizes an isolation transformer m the output stage of a two-stage device, Fig 2 is a preferred form of an electronic ballast in accordance with the invention which uses a SEPIC converter input state with a single integrated transformer- inductor, and Fιgs.3A-3H show waveforms useful for a better understanding of the present invention. Fig 1 illustrates a general pnor art high frequency electronic ballast circuit for operating one or more gas discharge lamps Rip A 50 or 60 Hz source of AC supply voltage 1 is connected to the input of an EMI filter consisting of a pair of magnetically coupled inductors LO and a capacitor CO. The output of the EMI filter is connected to a pair of input terminals of a 4 diode full wave bπdge rectifier 2 A first DC output terminal of the bπdge rectifier is connected to one terminal of a boost inductor LI which is part of a transformer 3 The second bndge rectifier output terminal is connected to a common line 4 The other terminal of inductor LI is connected to a common junction point 5 between a diode D5 and a transistor power switch Ql A current sensing resistor 6 is connected in senes circuit with the transistor power switch Ql to the common line 4 The junction point 12 of transistor switch Ql and the sensing resistor 6 is connected as a first control input to a control circuit 7, for example an integrated circuit manufactured by Motorola Corporation and designated MC34262 This control circuit is descnbed in a technical data publication by Motorola Corporation published in 1993 The control circuit has an output line 8 that controls the on/off switching of transistor switch Ql The diode D5 is connected in seπes circuit with a storage capacitor Cl across the seπes circuit of transistor power switch Ql and sensing resistor 6 An output stage is provided with a half-bndge inverter including transistor power switches Q2 and Q3 connected in senes circuit with a further current sensing resistor 9 across the storage capacitor Cl A blocking capacitor C2 and a resonant inductor L2 are connected in seπes between a junction point 10 between transistor switches Q2 and Q3 and one terminal of the pπmary winding of an output isolation transformer Tl The other terminal of the transformer pnmary winding is connected to the common line 4 A resonant capacitor C3 is connected in parallel with the output transformer pπmary winding A control input line is coupled to a junction point between resonant inductor L2, resonant capacitor C3 and the one terminal of the pnmary winding and to a first control input terminal of a second control circuit 11 which has two output control lines coupled to respective control electrodes of switching transistors Q2 and Q3 A second control line couples the voltage developed across sensing resistor 9 to a second control input of the high voltage control circuit 11, for example, the integrated circuit UBA 2010 A third control line connects the junction point 10 to a third input of the control circuit 11 A high voltage control IC suitable for use as the control circuit 11 is descnbed in UBA2010 Specification sheet by Philips The secondary winding of output transformer Tl is connected to a bank of four parallel connected discharge lamps Rip via four respective senes connected ballast capacitors Clp The transistor switch Ql is penodically turned on and off by control signals delivered to its control electrode from control circuit 7 via the output control line 8 The control circuit 7 switches under the control of signals supplied by the secondary winding of boost inductor LI, the voltage on storage capacitor Cl and a signal determined by the current flow through transistor switch Ql The input to the front end boost converter is a full wave rectified sinusoidal input line voltage at 50 Hz or 60 Hz When the power switch Ql is off the diode D5 is turned on and carnes a current for storage capacitor Cl based upon the electπc energy stored in the boost inductor LI The voltage stored on capacitor Cl provides the operating voltage for the voltage fed half-bπdge inverter including power switches Q2 and Q3 Inductor L2 and capacitor C3 form a resonant circuit at the switching frequency of the half-bπdge inverter The operation of this high frequency electronic ballast circuit is well known and will therefore not be descnbed in further detail A preferred embodiment of the invention is shown in Fig. 2. A low frequency source of AC supply voltage 1, e.g. 50 Hz or 60 Hz, is connected to the input of an EMI filter consisting of a pair of magne ically coupled inductors L0 and a capacitor CO. The output of the EMI filter is connected to a pair of input terminals of a four diode full wave bπdge rectifier 2. This part of the inverter is similar to Fig. 1 above. The high power electronic ballast circuit of Fig. 2 features a single integrated transformer-inductor combination Tl-13. This transformer-inductor combination also includes a control winding 14. The transformer Tl pnmary winding, the inductor 13 and the control winding 14 are all mounted on a common magnetic core, as indicated schematically by the dashed line 15. One possible implementation of the transformer-inductor combination is shown m Fιg.4, which comprses a pair of magnetic c cores in facing relationship with an air gap of length h therebetween. The pnmary and secondary windings of the transformer Tl are closely wound together on the left hand legs of the c cores. The inductor winding 13 and the control winding 14 are wound in the nght hand core legs as shown A seπes circuit including the inductor 13 (LI), a capacitor C5 and the isolation transformer pnmary winding L2 is connected to first and second DC output terminals of the full wave bndge rectifier 2. A common junction point 16 between inductor 13 and capacitor C5 is connected via a second senes circuit consisting of a transistor power switch Ql and a sensing resistor 6 to the common line 4 The control winding 14 of the transformer-mductor is connected to a first control input of a control circuit 7 and to ground. The junction point 12 between transistor switch Ql and sensing resistor 6 is connected to a second control input of the control cπcuit 7 and the first DC output of the bπdge rectifier circuit 2 is connected to a third control input of the control circuit 7. An output line 8 of the control circuit is connected to the control electrode (gate) of the switching transistor Ql The control circuit 7 may again utilize the Motorola IC, MC34262 The secondary winding L3 of isolation transformer Tl is connected to the anode of diode D5 and to a further common line 16 The cathode of diode D5 is connected to one terminal of storage capacitor Cl, whose other terminal is connected to the common line 16. Switching power transistors Q2 and Q3 of the half-bπdge inverter are connected in seπes circuit with sensing resistor 9 between the cathode of diode D5 and the common line 16. A blocking capacitor C2 is connected in seπes circuit with a resonant circuit including inductor L4 and capacitor C3 between the junction point 10 between switching transistors Q2 and Q3 and the common line 16 The junction point 10 is also connected to a control input of the high voltage control IC 11. The IC 11 is similar to the corresponding control circuit 11 of the apparatus of Fig 1 A second control input of the high voltage control circuit 11 is connected to a junction point of the resonant inductor L2 and the resonant capacitor C3 so that this control input is responsive to the lamp voltage The third control input of control circuit 11 is connected to the junction point of transistor switch Q3 and the sensing resistor 9 so that it receives a control signal determined by the current flow in the transistor switch Q3 A parallel connected bank of four discharge lamps Rip is connected in seπes with respective ballast capacitors Clp between the common line 16 and the junction point between the resonant inductor L2 and the resonant capacitor C3 The SEPIC converter in Fig 2 is operated in the critical conduction mode in order to provide good power factor correction In contrast to the traditional control schemes such as a constant operating frequency with either a continuous conduction mode (CCM) or a discontinuous conduction mode (DCM), the cπtical conduction mode of operation descnbed herein provides the advantage of a better tradeoff between the conflicting requirements of a simple control circuit and low circuit losses As mentioned above, in the CCM mode the control is more complex although the circuit losses are relatively low, whereas in the DCM operation mode the control is simpler but the circuit losses are higher The input voltage, Vin, at the output of the bndge rectifier 2 is a full wave fluctuating replica of the 50 Hz or 60 Hz sinusoidal AC voltage of the supply voltage source 1 The control circuit 7 switches the transistor switch Ql at a high frequency, for example, 50 KHz, so that for all intents and purposes for each cycle of this high switching frequency the low frequency input voltage, Vm, appears as a constant voltage as there is only a minute voltage change of Vin, at 50 Hz or 60 Hz, in one cycle of the 50 KHz frequency Fig 3 A shows the waveform of current flow (IQ1) through the transistor switch Ql as a function of time When this transistoi switch is turned on, the current IQ1 begins to ramp up from its initial value of zero A current IL1 flows through inductor 13 and also begins to ramp up from its zero value, as shown by the waveform in Fig 3B The transistor switch Ql continues to conduct dunng the peπod to to ti Dunng this time peπod, a nearly constant voltage VL2, see Fig 3C, appears across the pnmary winding of isolation transformer Tl and a current therein IL2, Fig 3D, ramps up from its zero value In this time penod, a current IC5, Fig 3E, flows through capacitor C5 At this time, diode D5 is off, see Fig 3F, and there is no current flow through this diode, see Fig 3G, current ID5 Therefore, there is no current flow, IC1, to capacitor Cl, see Fig 3H At the time instant tl, the voltage at the left-hand terminal of inductor 13 together with the voltage across the sense resistor 6 signal the control circuit 7 to turn off the transistor Ql. The current IQ1 in transistor Ql drops rapidly to zero, the current IL1 through the inductor 13 begins to ramp down towards the zero value, the transformer primary voltage VL2, Fig. 3C, reverses polarity, whereupon the primary current IL2, Fig. 3D, begins to ramp down, the diode D5 now turns on and a diode current ED5, Fig. 3G, begins to flow. As a result, a charge current IC1 flows into capacitor Cl as shown in Fig. 3H. At the time t 2 , the current IL2 through inductor 13 or the transformer Tl primary winding becomes zero and via the control winding 14 of the transformer, the control circuit 7 receives a turn-on signal and immediately produces a turn-on signal for transistor switch Ql over gate control line 8. The current IL1 in inductor 13 is also zero at this point in time, see Fig. 3B. The current IQ1 through transistor switch Ql now begins to ramp up again and a new cycle as described above repeats itself. The diode D5 is again cut off and the only current IC1 that flows through capacitor Cl is that required to operate the half bridge inverter in the output stage and the discharge lamps energized thereby. The output voltage is essentially constant except for a very small ripple voltage. The operation of the half-bridge resonant inverter output stage is conventional and need not be described in detail. The high voltage control circuit 11 alternately switches the transistor switches Q2 and Q3 on and off periodically as a function of control signals derived across the resonant capacitor C3 and the current sensing resistor 9. In the electronic ballast of Fig. 2, due to the rectified sinusoidal input line voltage Vin, the input output voltage gain, defined as Vo/Vin, is changed with time. In order to produce a sinusoidal input current in phase with the input line voltage, the SEPIC converter is operated in the critical conduction mode with a variable frequency during the input line frequency cycle. This operating mode is maintained by the PFC control IC by means of the voltage sense signal Vs developed in the transformer control winding 14. In the high frequency operating interval, the power switch Ql is turned on when the voltage sense signal Vs in control winding 14 reaches zero. With the turn on of switch Ql, the input line voltage Vin is applied to inductor 13. At the same time, the voltage across C 5 , equal to V in , is applied to the primary winding of transformer Tl. The current in the magnetizing inductance is increased as shown in Fig. 3. The current in transistor Ql increases approximately linearly (see Fig. 3A). At the same time, the energy storing capacitor C5 is discharged through transistor switch Ql and the primary winding of transformer Tl with a current IC5 as shown in Fig. 3C. When the current peak of IQ1, with proper scaling of sensing resistor 6, reaches the level of a reference curve of the input voltage Vin at time ti, the transistor switch Ql is turned off. The circuit then enters its second operating interval. In this interval, diode D5 is on and conducts current from transformer Tl to the output capacitor Cl. The current waveform ED5 of diode D5 is shown in Fig. 3G. In this interval, t]-t 2 , the voltage across transformer Tl leads to a gradual reduction of the inductor current IL1 as shown in Fig. 3B. When the current IL1 reaches zero, control winding 14 senses the signal (Vs) and triggers on transistor switch Ql to start the next switching cycle. The waveforms illustrated in Fig. 3 are derived only at a particular operating point during the low frequency line cycle. At other operating points, the shapes of the waveforms are the same, but with a different frequency and amplitude. In comparison with the prior art circuit shown in Fig. 1, the invention in accordance with Fig. 2 uses a single integrated transformer inductor in the input stage rather than a separate inductor in the input stage and a separate isolation transformer in the output stage. The electronic ballast according to the invention has a reduced number of magnetic components resulting in a reduction in size and cost of the electronic ballast. As compared to the electronic ballast of U.S. Patent 5,084,653, the invention of Fig. 2 utilizes magnetic components of reduced size and therefore reduced cost. Although a preferred embodiment of the present invention has been shown and described herein, it will be obvious that such embodiment is provided by way of example only. Numerous variations, changes and substitutions will occur to those skilled in the art without departing from the spirit and scope of the invention. Accordingly, it is intended that the dependent claims cover all such embodiments as fall within the spirit of the invention.

Claims: CLAIMS: 1. A high frequency electronic ballast circuit for one or more discharge lamps comprising: first and second input terminals for connection to a source of low frequency supply voltage (1) for the electronic ballast circuit, a rectifier coupled to the input terminals for rectifying the low frequency supply voltage, an input stage coupled to output terminals of the rectifier and comprising; a transistor switch (Ql), a transformer-inductor (Tl,13) having a primary winding, a secondary winding and an inductor (13), a capacitor (C5), a rectifier (D5), a storage capacitor (Cl), first means connecting the inductor (13), the capacitor (C5), the transformer (Tl) primary winding and the transistor switch (Ql) to said output terminals of the rectifier to form a high-frequency converter circuit, a control circuit (7) supplying a high frequency switching signal to the transistor switch as a function of at least first and second control signals derived from said transformer-inductor, second means connecting the rectifier (D5) and storage capacitor (Cl) to the transformer (Tl) secondary winding, and an output stage comprising; a DC/AC inverter circuit including an LC (L4-C3) resonant circuit, and connection terminals for at least one discharge lamp (Rip). 2. The high frequency electronic ballast circuit as claimed in claim 1 wherein the control circuit (7) operates said input stage in the critical conduction mode of operation. 3. The high frequency electronic ballast circuit as claimed in claim 2 wherein the transformer-inductor further comprises a control winding (14) which supplies said first control signal to a first control input of the control circuit (7) whereby the control circuit responds thereto so as to turn on the transistor switch (Ql) when the current in the inductor (13) reaches a zero value. 4. The high frequency electronic ballast circuit as claimed in claim 1 wherein the control circuit (7) receives said first control signal at a first control input thereof and the control circuit responds thereto so as to turn on the transistor switch (Ql) when the current in the inductor (13) reaches a zero value. 5. The high frequency electronic ballast circuit as claimed in claim 1 wherein the control circuit (7) is further responsive to a third control signal (at 12) determined by a current flowing through the transistor switch for deriving the high frequency switching signal for the transistor switch. 6. The high frequency electronic ballast as claimed in 1 wherein the first connecting means connects the inductor (13), the capacitor (C5) and the transformer (Tl) primary winding in series circuit between the output terminals of the rectifier and connects the transistor switch (Ql) to a junction point (16) between the inductor (13) and the capacitor (C5) and to an output terminal of the rectifier. 7. The high frequency electronic ballast circuit as claimed in claim 1 wherein the transformer-inductor (13,T1,14) comprises a single integrated magnetic component including the inductor (13), the transformer (Tl) primary and secondary winding, and a control winding (14) for the control circuit (7), all on a common magnetic core (15) means. 8. The high frequency electronic ballast circuit as claimed in claim 7 wherein the DC/AC inverter circuit comprises; second (Q2) and third (Q3) transistor switches connected in series across the storage capacitor (Cl), wherein the LC resonant circuit includes a resonant inductor (L2) and a resonant capacitor (C3) serially connected across the third (Q3) transistor switch, and a high voltage control circuit (11) for controlling the switching of the second and third transistors in response to third and fourth control signals determined by the voltage across the resonant capacitor (C3) and the current flow through the third transistor switch (Q3-via 9). 9. The high frequency electronic ballast circuit as claimed in claim 1 wherein the first and second input terminals are adapted to supply a low frequency full wave rectified sinusoidal-type supply current for the electronic ballast circuit, and the inductor (13) supplies said second control signal to a second control input of the control circuit (7) whereby the control circuit triggers the transistor switch (Ql) into cut-off when the current flow in the inductor is equal in amplitude to the amplitude of a low frequency supply current at the input terminals.

Inventors: CHANG CHIN, HERNANDEZ ADAN, CHANG, CHIN, HERNANDEZ, ADAN
Applicants: KONINKL PHILIPS ELECTRONICS NV, KONINKLIJKE PHILIPS ELECTRONICS N.V.
Application year: 2000
Main classification: H05B41
IPC codes: H05B41