Method of constructing a space-time constellation

Abstract: Disclosed is a method for wireless signal transmission of signals from an array of two or more antennas, in which each signal to be transmitted is selected from a constellation of unitary space-time signals. Each unitary space-time signal is a unitary matrix, in which each column represents a respective antenna, each row represents a respective time interval, and each element represents a complex amplitude to be transmitted by a given antenna during a given time interval. In specific embodiments of the invention, the matrices of the signal constellation form a non-Abelian group which has a positive diversity product, or a set derived from such a group. In other embodiments, the signal constellation is a subset of such a group, and its multiplicative closure forms a finite non-Abelian group which has a positive diversity product. In still other embodiments, the signal constellation is an extension of any of the preceding types of constellations, formed by adding one or more further elements that do not belong to, and are not derived from, the group or group subset.

Description: This invention relates to methods of constructing a space-time constellation. When wireless communication signals are in transit between a transmit antenna and a receive antenna, they are generally subject to destructive interference and other physical effects that vary in time. As a consequence, the received signal arrives with an attenuation and phase delay that will also generally vary in time. This effect is referred to as fading. Where, e.g., attention is confined to a sufficiently narrow bandwidth, the attenuation and phase delay of the received signal can be described by a complex-valued coefficient h often referred to as the fading coefficient. Practitioners in the field of wireless communications have recognized that by using multiple antennas for transmission, reception, or both, it is possible to mitigate some of the undesirable consequences of fading, and to achieve certain other benefits as well. For example, the use of multiple antennas affords alternate transmission paths, some of which may, at a given time, be less subject to fading than others. The use of multiple antennas also provides a mechanism for sending redundant signals, the better to understand them at the receiving end, even in the face of severe fading. Even if redundancy is not a primary objective, the use of multiple antennas can provide a mechanism for increasing total transmission rates in a given frequency channel, by simultaneously transmitting multiple, independent signals that can be separated at the receiving end. FIG. 1 is a simplified, schematic diagram of a wireless communication system having two transmit antennas 10, 15, and three receive antennas 20, 25, 30. As indicated at block 40, baseband-level signals generated at block 35 are modulated onto a carrier wave, which is shown symbolically in the figure as generated at oscillator 50. It will be seen from the figure that each of the receive antennas receives transmission from, e.g., antenna 10. Provided there is sufficient spatial separation, for example spatial separation, between the receive antennas, the transmitted signals will bear distinct fading effects when they are received. (In this regard, it should be noted that diversity of fading effects can in at least some cases also be achieved by using receive antennas that are selectively receptive to diverse polarizations of the incoming signal, even if the antennas are not substantially separated in space.) A separate fading coefficient h mn accounts for the fading effects, in the physical propagation channel, between each transmit antenna m and each of the receive antennas n. As shown in the figure, m = 1 for antenna 10 and m = 2 for antenna 15. Similarly, n = 1 for antenna 20, n = 2 for antenna 25, and n = 3 for antenna 30. All six of the fading coefficients h mn are arranged in a matrix H, denoted by block 55 of the figure. It will be clear from the foregoing that each receive antenna receives, during a given time interval, a total signal that is a weighted sum of the transmissions from the respective transmit antennas. The weight coefficients of that sum are the fading coefficients. The received signal is also typically corrupted by additive noise, which is not indicated in the figure. Because each of the receive antennas is typically receiving a different weighted sum of the transmitted signal, it is theoretically possible under certain conditions to recover the transmitted baseband-level signals by taking appropriate weighted combinations of the demodulated, received signals. One condition necessary for such recovery is that there must be at least as many receive antennas as there are transmit antennas. Another such condition is that the additive noise must not be excessive relative to the signal strength. (It should be noted in this regard that practical methods of signal recovery often imply indirect methods such as maximum-likelihood detection, which is described below.) Turning again to FIG. 1, demodulation of the received signals is indicated at blocks 60-70. Signal recovery is indicated at block 75. The recovered signals are indicated at block 80. A new kind of transmitted signal, referred to as a space-time signal has been shown to offer potential improvements in both fading performance and transmission rate. In space-time modulation, each signal that is sent is selected from a finite set, or constellation , of predetermined signal matrices. Thus, if there are a total of L such matrices, each individual matrix that is transmitted conveys log 2 L bits of information. Advantageously, the signal matrices are unitary . In a unitary matrix, the columns or rows (whichever are longer) are mutually orthogonal and have unit norm. The individual elements of a unitary matrix are complex numbers; i.e., numbers that are real, imaginary, or sums of real and imaginary components. When unitary matrices are used, the modulation method is referred to as unitary space-time modulation (USTM). The method of transmitting space-time matrices will now be explained with reference to FIG. 1. A 2 × 2 space-time signal matrix is represented in block 35 as the matrix Each element of this matrix is a complex number. Such an element is for example modulated onto the carrier wave by subjecting the carrier wave to a suitable pulse-shaping function having the corresponding complex amplitude and having a width appropriate to the length of a transmission time interval. Each such transmission time interval is referred to as a channel use. During the first channel use, element s 11 is transmitted by antenna 10 and element s 12 is transmitted by antenna 15. During the second channel use, element s 21 is transmitted by antenna 10, and element s 22 is transmitted by antenna 15. More generally, each row of a space-time matrix corresponds to a respective channel use, and each column corresponds to a respective transmit antenna. Thus, the entry in row p and column q is the complex amplitude transmitted during the p 'th channel use by the q 'th antenna. The length of a channel use is generally chosen to be no longer than a fading interval; i.e., a length of time over which the fading coefficients can be assumed constant. At the receiving end, the transmitted signal matrix is recovered as the matrix as indicated at block 80 of FIG. 1. There are various methods for recovering an estimate of the transmitted signal matrix, based on the received matrix. According to one such method, referred to as maximum likelihood (ML) detection, a likelihood score is computed for each candidate signal matrix that might have been transmitted, and that candidate which maximizes the likelihood score is identified as the transmitted matrix. Typically, the likelihood score is the probability of the raw, signal amplitudes that are received at baseband level, given that the candidate matrix was transmitted. When each signal matrix is transmitted as a matrix drawn from the signal constellation, the ML detector must typically be provided with values of the fading coefficients. Such values may, for example, be measured using appropriate pilot signals. However, there are alternative transmission techniques, referred to as unknown channel techniques, that do not require the ML detector to know the (approximate) fading coefficients, provided the fading coefficients do not change substantially over at least as many, typically at least twice as many, channel uses as there are transmit antennas. One class of unknown channel techniques is referred to as differential modulation. An example of differential modulation is described in greater detail below. Briefly, each signal to be transmitted is the product of the previously transmitted signal, times a new signal matrix selected from the signal constellation. In that case, a suitably adapted ML detector can take advantage of the commonality of fading effects between each signal and its predecessor to effectuate signal recovery without explicit knowledge of the fading coefficients. One relatively early study of space-time signals based on trellis codes, for transmission over a known channel, is described in V. Tarokh, et al., "Space-time codes for high data rate wireless communication: Performance criterion and code construction," IEEE Trans. Info. Theory 44 (1998) 744-765. Unitary space-time modulation, applicable to both known and unknown channels, is described, e.g., in the U.S. Patent Application 6327310, filed on August 14, 1998 by B.M. Hochwald et al. under the title, "Wireless Transmission Method for Antenna Arrays, Having Improved Resistance to Fading," the U.S. Patent Application 6363121, filed on December 7, 1998 by B. Hochwald et al. under the title, "Wireless Transmission Method for Antenna Arrays Using Unitary Space-Time Signals," and the co-pending U.S. Patent Application Serial No. 09/528973, filed on March 21, 2000 by B. Hassibi et al. under the title, "Method of Wireless Communication Using Structured Unitary Space-Time Signal Constellations," all commonly assigned herewith. To help make unknown-channel multiple-antenna communication practical, a differential multiple-antenna modulation scheme using differential unitary space-time signals has been proposed that is well-tailored for unknown continuously varying Rayleigh flat-fading channels. Differential unitary space-time signals are unitary matrix-valued signals that are a multiple-antenna generalization of the standard differential phase-shift keying (DPSK) signals commonly used with a single antenna over an unknown channel. Differential unitary space-time modulation schemes are described, e.g., in the co-pending U.S. Patent Application Serial No. 09/356387 (family member of EP 1069723), filed on July 16, 1999 by B. Hochwald et al. under the title, "Method for Wireless Differential Communication Using Multiple Transmitter Antennas," commonly assigned herewith. Bell Laboratories Technical Memorandum (01-1999); Hochwald et al., Systematic Design of Unitary Space-Time Constellations describes signal constellations generated from a predetermined generator matrix and a matrix, which in some sense may be regarded as a rotation matrix, which is an L'th root of the identity matrix. A signal constellation is generated by applying each of the L distinct powers of the rotation matrix to the generator matrix. Significantly, the set of all powers of the rotation matrix constitutes a group which is cyclic, and hence Abelian. The resulting signal constellation does not necessarily constitute a group. The same Bell Laboratories Technical Memorandum also describes signal constellations generated by applying products of powers of a finite set of diagonal matrices to a generator matrix. The diagonal matrices are all q'th roots of the identity matrix for some integer q, and together they generate an Abelian group. There remains a need for a principled approach to designing space-time signal constellations that will more fully achieve their theoretical benefits of high data rate and low error probability. According to this invention there is provided a method as claimed in claim 1. According to our new approach, signal constellations of unitary space-time matrices are constructed to have group properties under the operation of matrix multiplication. A set G is a group under a binary multiplication operation if it is closed under multiplication, satisfies the associative law, has an identity element, and contains a multiplicative inverse for each element. As noted above, the matrix that is detected at the receiving end of the communication system bears the effects of fading and of additive noise. As a consequence, there is some likelihood of error, at the receiver, in ascribing a given received signal to a particular transmitted signal. One important feature of a space-time signal constellation is that each pair of signal matrices should differ so distinctly from each other that they are unlikely to be confused by the receiver even in the presence of fading and noise. We have found that signal constellations derived from certain types of groups have a relatively high diversity product. The diversity product is a measure, discussed in more detail below, of how distinguishable the signals are after transmission over a fading channel. Thus, such signal constellations are particularly advantageous for use in space-time modulation. In one broad aspect, our invention involves a method for wireless signal transmission of signals, in which each signal to be transmitted is selected from a constellation of unitary space-time signals. In some embodiments of the invention, the signal constellation forms a group. In other embodiments, the signal constellation does not contain every member of a group, but a group, referred to as a multiplicative closure, is formed by the set of all possible products of the members of the signal constellation. In still other embodiments, the signal constellation is derived from a group or from a subset of the group by multiplying each member of the group or group subset by a common element, which will typically be a unitary matrix. The resulting constellation is said to be a coset of the group or group subset. Significantly, the group in each of these cases is a non-Abelian group of matrices having a positive diversity product. A group is non-Abelian if it contains at least two elements A, B, for which AB≠BA. Brief Description of the Drawing FIG. 1 is a schematic block diagram of a system for wireless communication using multiple-antenna arrays. FIG. 2 is a graph of the theoretical block-error rate performance of a signal constellation based on the group SL 2 (F 5 ), in one embodiment. The antenna configuration consists of M = 2 transmitter antenas and N = 1 receiver antenna. Included in the figure, for comparison, are theoretical results for the same antenna configuration but in which the signal constellation is based on a non-group orthogonal construction (dashed line), the best diagonal construction, which was based on an Abelian group (dashed-dotted line), and a quatemion group (dotted line). FIG. 3 is a graph of the theoretical block-error rate performances as in FIG. 2, but with N = 2 receiver antennas. This figure suggests that the coding advantage of the group SL 2 (F 5 ) becomes more pronounced as the number of receiver antennas increases. FIG. 4 is a graph of the theoretical block-error rate performance of a signal constellation based on the group F 15,1,11 , in one embodiment. The antenna configuration consists of M = 2 transmitter antennas and N = 1 receiver antenna. The solid line represents F 15,1,11 , which has L = 240 unitary matrices (the rate R = 3.95). The dashed line represents a non-group orthogonal design ( R = 4). The dashed-dotted line represents a diagonal (Abelian group) construction ( R = 3.95). The dotted line represents a quaternion group with L = 256 matrices ( R = 4). FIG. 5 is a graph of the theoretical block-error rate performance of a signal constellation based on the group G 21,4 , which has an irreducible representation of L = 63 matrices. The antenna configuration consists of M = 3 transmitter antennas and N = 1 receiver antenna ( R = 1.99). For comparison, we have also represented in the figure a diagonal (Abelian group) construction having the same rate R . FIG. 6 is a graph of the experimentally measured block-error rate performance of a signal constellation based on the group G 21,4 (as in Figure 5) transmitted over wireless apparatus in an indoor hallway of our research center. FIG. 7 is a graph of the theoretical block-error rate performance of a signal constellation based on the group K 1,1,-1 (solid line), which is the binary extension of the group SL 2 ( F 5 ) having L = 240 unitary matrices ( R = 1.98). The antenna configuration consists of M = 4 transmitter antennas and N = 1 receiver antenna. For comparison, we have also represented in the figure the best diagonal (Abelian group) construction found for that antenna configuration and rate (dashed line). Detailed Description The design problem for unitary space time constellations is: let M be the number of transmitter antennas and R the desired transmission rate (in bits/channel use); construct a set ν of L = 2 RM unitary M x M matrices such that for any two distinct elements A and B in ν, the quantity |det ( A-B )| is as large as possible. Any set ν such that |det ( A-B )| >0 for all distinct A,B ∈ ν is said to have full diversity. The design problem can advantageously be simplified by imposing, a priori , an appropriate structure on the constellation set ν. Here, we focus on sets ν of unitary matrices that form a mathematical group with respect to matrix multiplication. The use of a group structure offers certain advantages. The first, based on theoretical considerations, is that constellations having group structure are generally expected to have good potential performance. The second advantage is based on practical considerations. Since in differential space-time modulation one multiplies matrices in ν to form the transmitted signal matrix, if ν is a group, every transmitted signal matrix is always an element of ν. Therefore, explicit matrix multiplication is replaced by the simpler group table-lookup. Some of the groups that emerge as good signal sets are rather surprising. We have found, for example, that if M is odd, there is only a single class of possible groups. If M = 2 or M = 4, some of the signal sets that are excellent performers involve SL 2 ( F 5 ) - the special linear group in two dimensions over the field F 5 , i.e., over the field of integers modulo 5 with the binary operations of addition and multiplication. However, we also found that full-diversity non-Abelian groups do not necessarily exist for all M and R . The Ravleigh flat fading channel. Consider a communication link with M transmitter antennas and N receiver antennas operating in a Rayleigh flat-fading environment. The nth receiver antenna responds to the symbol sent on the mth transmitter antenna through a statistically independent multiplicative complex-Gaussian fading coefficient h mn . The received signal at the nth antenna is corrupted at time t by additive complex-Gaussian noise w tn that is statistically independent among the receiver antennas and also independent from one symbol to the next. We assume that time is discrete, t = 0,1,.... It is convenient to group the symbols transmitted over the M antennas in blocks of M channel uses. We use τ= 0,1,... to index these blocks; within the τth block, t = τ M ,..., τM + M - 1 . The transmitted signal is written as an M × M matrix S τ whose m'th column contains the symbols transmitted on the m 'th antenna as a function of time; equivalently, the rows contain the symbols transmitted on the M antennas at any given time. The matrices are normalized so that the expected square Euclidean norm of each row is equal to one. Hence, the total transmitted power does not depend on the number of antennas. The fading coefficients h mn are assumed to be constant over these M channel uses. Similarly, the received signals are organized in M × N matrices X τ . Since we have assumed that the fading coefficients are constant within the block of M symbols, the action of the channel is given by the simple matrix equation X τ = ρ S τ H τ + W τ for τ = 0,1,.... Here H τ = { h mn } and W τ = { w tn } are M × N matrices of independent CN (0,1)-distributed random variables; i.e., variables complex-normal distributed with zero mean and unit variance. Because of the power normalization, ρ is the expected SNR at each receiver antenna. Known channel modulation. We first discuss signal encoding and decoding when the receiver knows the channel H τ . We assume-that the data to be transmitted is a sequence z 0 , z 1 , ... with z τ ∈ {0,..., L -1}. The data then simply dictate which matrix is transmitted: S τ = V z r , wherein V zr is a unitary data matrix taken from the constellation for transmission at time τ. Each transmitted matrix occupies M time samples of the channel, implying that transmitting at a rate of R bits per channel use requires a constellation ν = { V 1 ,..., V L } of L = 2 RM unitary signal matrices. The receiver knows H τ and computes the maximum likelihood estimate of the transmitted data as where the matrix norm is the Frobenius norm || A || 2 = tr ( A + A ) . The quality of a constellation ν is determined by the probability of mistaking one symbol of ν for another. The Chemoff bound on the pairwise probability of mistaking ν l for V l , with a known channel (averaged over the statistics of H) is given by where σ m ( V l - V l ' ) is the mth singular value of the M × M matrix V l - V l ' . Differential unitary space-time modulation. When the receiver does not know the channel, one can communicate using multiple-antenna differential modulation. Multiple-antenna differential modulation is formally similar to standard single-antenna differential phase-shift keying. In standard DPSK, the transmitted symbol has unit-modulus and is the product of the previously transmitted symbol and the current data symbol. The data symbol typically is one of L equally-spaced points on the complex unit circle. As a generalization, M-antenna differential unitary space-time modulation differentially encodes a M x M unitary matrix that is the product of the previously transmitted matrix and a unitary data matrix taken from the constellation. In other words, S τ = V z r S τ-1 , τ=1,2,..., with S 0 = I M . We immediately see why it is useful in practice to have ν form a group under matrix multiplication: if ν is a group then all the transmitted matrices S τ also belong to ν. Therefore, the transmitter sends matrices S τ from a finite set and does not need to explicitly multiply S τ = V z r S τ- 1 , but rather can use a group table-lookup. If the fading coefficients are approximately constant over 2 M time samples ( H τ ≈ H τ-1 ), the received matrices turn out to obey X τ = V z r X τ-1 + 2 W ' τ , where W ' τ is a M × N matrix of additive CN (0,1) noise. A known maximum likelihood decoder has the simple structure and the Chernoff bound on the pairwise probability of error with differential modulation on an unknown channel is At high signal-to-noise ratio (SNR), the bound on the pairwise probability of error, whether the channel is known or unknown, depends primarily on the product of the singular values, which is the modulus of the determinant of V l - V l ,. In other words, for high SNR we may write where α=4 when the channel is known and α=8 when the channel is unknown and used differentially. Therefore, we measure the quality of a constellation ν by its so-called diversity product The scaling 1 / 2factor guarantees that 0 ≤ ζν ≤ 1. The exponent 1 / M essentially gives the geometric mean of the M singular values since the modulus of the determinant is the product of the singular values. Clearly, a constellation with larger ζ ν is superior. Any constellation with ζν > 0 is said to have full diversity. When ζ ν > 0 and the SNR is high, we note that no two transmitted signals can give the same received signal X , for any H . Here, we consider only full-diversity constellations and, in particular, we try to find constellations with diversity product ζν as large as possible. Group representations We wish to find a set ν of L unitary matrices for which the diversity product ζ ν is as large as possible. In this section we constrain ν to form a group under matrix multiplication. With the group requirement, the design problem becomes that of finding a group of L unitary M × M -matrices such that is as large as possible. (The matrix symbol I d denotes the d × d -identity matrix. We later omit the dimension d if it is clear from the context.) Groups fall into many different classes distinguished, inter alia, by whether the binary multiplication operation is Abelian (i.e., commutative), and by regularities of various kinds that emerge in the group multiplication tables. One broad family of groups, encompassing groups of many different classes, is the family of non-Abelian fixed-point-free groups. One important property of fixed-point-free groups is that any such group has at least one representation with full diversity. A representation is a mapping to a group of invertible matrices, such that the product of two elements of the represented group is mapped to the product of the two corresponding image matrices, and such that the identity element of the group is mapped to the identity matrix. It should be noted that any finite group has a representation in terms of unitary matrices. A non-Abelian class of fixed-point-free groups. Let G m.r = 〈σ,τ|σ m = 1, τ n = σ t , σ τ σ τ 〉, where n is the order of r modulo m (i.e., n is the smallest positive integer such that r n ≡ 1 mod m ), t = m / gcd( r -1, m), and we have gcd (n,t) = 1. (We use the notation σ τ for σ,τ ∈ G to mean the element τστ -1 ) The greatest common divisor of r -1 and m is denoted r 0 . The group G m,r has order mn because it contains the subgroup 〈σ〉 of order m and index n (the term "index" refers to the number of cosets). G m , r is fixed-point-free if and only if all prime divisors of n divide gcd( r-1,m). We here limit our attention to groups in which n is greater than 1. This limitation excludes those groups of the class G m,r that are cyclic, as well as those that are Abelian. (A fixed-point-free group is Abelian if and only if it is cyclic.) We now present a few examples of the fixed-point-free groups G m , r . Example 1 (3 antennas). Let n = 3 and take r = 4 and m = 21. Then we have r 0 = 3, t = 7, gcd( n,t ) = gcd(3,7) = 1, and all prime divisors of n (i.e., the prime 3) divide r 0 . Hence, G 21,4 is a fixed-point-free group. If we set η = e 2π i /21 , and then the 63 matrices A l B k , l = 0,...,20, k = 0,1,2, form the group G 21,4 under matrix multiplication, with the identification σ ≡ A , τ ≡ B . We have ζ G 21.4 =0.3851. This 3-antenna, 63-element constellation is one element shy of having rate R =2. Example 2 (9 antennas). Let n = 9 and take r = 4 and m = 57. Then we have r 0 = 3 and t = 19, gcd( n,t ) = 1, and all prime divisors of n divide r 0 . Hence G 57,4 is fixed-point-free. If we set η = e 2πi /57 , and A= diag( η ,η 4 ,η 16 ,η 7 ,η 28 ,η 55 ,η 49 ,η 25 ,η 43 ), where diag ( a 1 ,..., a n ) denotes the diagonal matrix with diagonal entries a 1 ,..., a n , then the 513 matrices A l B k , where l = 0 ,..., 56, and k = 0 ,..., 8 form the group G 57,4 under matrix multiplication with the identification ρ≡ A , τ≡ B . We have ζ G 57.4 =0.361. This 9-antenna, 513-element constellation exceeds rate 1 by one element. Classification of fixed-point-free groups. We have classified all fixed-point-free groups and computed all the irreducible fixed-point-free representations of these groups. As will be appreciated by those skilled in the art, a representation Δ(G) of a group G having elements g is reducible if for some invertible complex linear transformation T and some pair Δ 1 (G), Δ 2 (G) of lower-dimensional representations, the following is true for all g ∈ G : If a group representation is not reducible, it is said to be irreducible. (The symbol " 0 " in the preceding expression represents a zero matrix of appropriate dimension.) Our results are presented below. The group types One type of fixed-point-free groups is G m,r for some m and r, presented above. There are five more types. For the following discussion, we introduce the following convention: Given a pair of integers (m,r), we implicitly define n to be the order of r modulo m ; we define r 0 = gcd( r -1, m ); and t = m / r 0 . We call the pair ( m , r ) admissible, if gcd (n,t) = 1, and all prime divisors of n divide r 0 . The six group types are: 1. G m,r (Described above): G m,r = 〈σ,τ|σ m = 1,τ n = σ t , σ τ = σ r 〉, where ( m,r ) is admissible. The order of G m , r is L=mn. As noted above, we here limit our attention to those groups of type G m , r for which n is greater than 1. 2. D m,r,l : D m,r,l = 〈σ,τ,γ|σ m = 1,τ n = σ t , σ τ = σ r , σ γ = σ l , τ γ = τ l ,γ 2 =τ nr 0/2 〉, where nr 0 is even, ( m,r ) is admissible, l 2 ≡ 1 mod m, l ≡ 1 mod n, and l ≡ -1 mod s , where s is the highest power of 2 dividing mn . The order of D m , r , l is L = 2mn. 3. E m.r : E m , r = 〈σ,τ, µ ,γ|σ m = 1,τ n =σ t , σ τ =ρ r ,µ σ mlt = µ , γ σ mlt =γ, µ 4 = 1, µ 2 = γ 2 , µ γ = µ - 1 ,µ τ =γ,γ τ = µγ〉 where (m,r) is admissible, mn is odd, and nr 0 is divisible by 3. The order of E m , r is 8 mn. 4. F m , r , l: F m , r , l = 〈σ,τ, µ ,γ,ν|σ m =1,τ n =σ t , σ τ =σ r ,µ σmlt =µ,γ σ mlt =γ , µ τ =γ,γ τ =µγ , µ 4 = 1,µ 2 = γ 2 , µ γ = µ -1 , ν 2 = µ 2 , σ v = σ l , τ v - τ l , µ v = γ -1 ,γ v = µ -1 〉 where ( m,r ) is admissible, mn is odd, r 0 is divisible by 3, n is not divisible by 3, l 2 ≡ 1 mod m, I ≡1 mod n , and l ≡ -1 mod 3. The order of F m , r , l is 16mn. 5. J m,r : J m , r = SL 2 (F 5 ) × G m , r , where ( m , r ) is admissible, gcd( mn ,120) = 1, and SL 2 ( F 5 ) is the group of 2 × 2-matrices over F 5 with determinant 1. SL 5 ( F 5 ) has the generators and relations SL 2 (F 5 ) = 〈 µ , γ | µ 2 = γ 3 = ( µ γ) 5 , µ 4 = 1〉. The order of J m,r is 120 mn . 6. K m , r , l : K m.r.l = 〈 J m , r , ν〉 with the relations ν 2 = µ 2 , µ ν = ( µ γ) 7 (γ µ ) 2 γ(γ µ ), γ ν = γ, σ ν =σ l ,τ ν = τ l , where µ and y are as in J m , r , and where l 2 ≡ 1 mod m, / ≡ 1 mod n. The order of K m , r , l is 240 mn Representation theorem. We have mathematically proven the following relationships between the above classes of fixed-point-free groups and their irreducible unitary representations: (1) G m , r for admissible (m,r) has an irreducible n -dimensional fixed-point-free representation given by and ξ = e 2π i / m . The corresponding constellation is given by the matrices A s B k , s = 0, ..., m-1 , k = 0 ,...., n-1. We note here (and omit in the remaining descriptions) that, implicitly, in this representation the matrix A becomes a scalar and B becomes undefined when r = 1 because G m, l is cyclic. (2) D m , r , l with admissible (m,r) has an irreducible 2 n -dimensional fixed-point-free representation given by where ξ = e 2 πi / m . The corresponding constellation is given by A s B k R j , s=0 ,..., m-1, k=0,..., n-1, j=0, 1 . (3) E m , r for admissible ( m , r ) has an irreducible 2 n -dimensional fixed-point-free representation given by where z = 1 if 9 divides m, and z = 3 otherwise. The corresponding constellation is given by A s / z B k / z P j Q P , s = 0 ,.... m -1 , k = 0 , ..., n - 1 , j = 0, ..., 3 , p=0 , 1. (4) If n > 1 or l ≢ 1 mod ( m /3), then F m , r , l with admissible ( m,r ) has an irreducible 4n-dimensional representation given by: where A z , B z , P, Q are the matrices defined for the group E m,r , and z = 1 if 9 divides m, and z = 3 otherwise. If r = 1 and I ≡ 1 mod ( m /3), then F m , 1 , l has an irreducible 2-dimensional fixed-point-free representation given by where A 0,3 , F 0 , and F 1 are the matrices defined for E m,r . The corresponding constellation is given by A s B k P j Q P R q , where s = 0 , ..., m - 1 , k = 0, ... , n - 1 , j = 0 , ..., 3 , p=0 , 1 , q=0 , 1. (5) J m,r has an irreducible 2n-dimensional fixed-point-free representation given by where η = e 2 πi /5 , ξ = e 2π i / m , and ⊗ denotes Kronecker-product. The corresponding constellation consists of the matrices A s B k ( PQ ) j X , s = 0, ... , m - 1 , k = 0, ..., n - 1., j =0, ..., 9, and X runs over the set {I 2 n , P , Q, QP , QPQ , QPQP , QPQ 2 , QPQPQ, QPQPQ 2 , QPQPQ 2 P , QPQPQ 2 PQ , QPQPQ 2 PQP}. (6) K m,r,l has an irreducible 4 n -dimensional fixed-point-free representation given by where η = e 2π i /5 , ζ = e 2π i / m , and ⊗ denotes Kronecker-product. The corresponding constellation is given by A s B k ( PQ ) j XR P , s = 0 , ..., m - 1 , k = 0 ,..., n - 1 , j = 0 , ..., 9, p = 0, 1, and X runs over the set { I 4n , P , Q, QP , QPQ , QPQP , QPQ 2 . QPQPQ , QPQPQ 2 , QPQPQ 2 P , QPQPQ 2 PQ , QPQPQ 2 PQP}. The six types of fixed-point-free groups are summarized in Table 1. For each group G represented in the Table, L is the order of G (the size of the constellation) and M is the dimension of the representation of G (number of transmitter antennas). Group Type L M Comments 1. G m , r mn n 2. D m , r , l 2mn 2n 3. E m , r 8mn 2n 4. F m , r , l 16 mn 4n if n > 1 or l ≢ 1 mod m /3 F m , l , l 16mn 2 if l ≡ 1 mod m /3 5. J m , r 120 mn 2 n 6. K m , r , l 240 mn 4n Some explicit simple constellations. Simple examples of some of the classes of fixed-point-free groups are presented below. For reasons of simplicity, we will identify the groups with images of their fixed-point-free representations computed above, and list the group elements as matrices. We start with groups that have an irreducible fixed-point-free representation for M = 2 transmitter antennas. 1. The smallest example of a G m , r having a 2-dimensional irreducible fixed-point-free representation is G 6 ,- 1 . The corresponding constellation consists of the 12 matrices A s B k , s = 0 , ..., 5, k = 0, 1, where and ξ = e 2πi/6 . Its rate is R = log (12)/2 = 1.79, and its diversity product is ζ G 6,-1 = 0.5. 2. The smallest example of a D m,r.l having a 2-dimensional irreducible fixed-point-free representation is D 4,1,-1 . The corresponding constellation consists of the 12 matrices A s B k , s = 0,...,5, k = 0,1, where We have ζ Q 2 = 2 /2 ≈ 0.7071. It should be noted that D 4,1,-1 is a special case of the so-called quaternion groups D 2 n ,1-1 , which were proposed for use in wireless communication in B.L. Hughes, "Differential Spacetime Modulation," Proc. IEEE Wireless Networking and Communications Conference (WCNC), vol. 1 , pp. 145-149 (New Orleans, Sept. 1999). 3. The smallest example of a group E m , l is the group E 3 , 1 of order 24. This group is isomorphic to SL 2 ( F 3 ), the group of 2 x 2 matrices over F 3 with determinant 1. The constellation is given by the 24 matrices A s P j Q p , where s = 0, 1, 2, j =0,...,3, p =0,1, and Its rate is R = 2.29, and ζ E 3,1 = 0.5. 4. The smallest example of a group F m,1,l is the group F 3,1,-1 which has 48 elements. It consists of the matrices A s P j Q p R q , where s = 0, 1, 2, j =0,..., 3, p = 0, 1, q = 0.1, and A, P, Q are as above while Because n = 1, the matrix B does not appear. The constellation has rate R = 2.79, and ζF 3.1-1 = 2- 2/ 2 ≈ 0.3868. 5. The smallest example of J m , r is J 1,1 which is isomorphic to SL 2 ( F 5 ). This constellation has 120 elements given by the matrices ( PQ ) j X , where j =0,..., 9, X runs over the set [ I 2 , P , Q , QP , QPQ , QPQP , QPQ 2 , QPQPQ, QPQPQ 2 , QPQPQ 2 P , QPQPQ 2 PQ , QPQPQ 2 PQP }, and where η = e 2 πi /5 . It has rate R = 3.45, and ζ SL 2 (F 5 ) = 1 2 (3- 5 )/2 ≈ 0.3090. This group performs especially well. 6. The simplest example of a fixed-point-free group with irreducible fixed-point-free representations for M = 3 is the group G 21,4 described above. 7. The best example of a low-rate fixed-point-free group with an irreducible 4-dimensional fixed-point-free representation is K 1,1,-1 . It has 240 elements, with rate R = log 2 (24)/4 = 1.98, and ζ K 1,-1,-1 = 0.5. The elements of this constellation are given by ( PQ ) j XR p , where j =0,...,9, p = 0,1, X runs over the same set as in example (5), above, but with Exemplary constellations and their performance. We will now discuss the performance of some of the constellations above. To evaluate the performance, we used the differential transmission framework described above. Most of the constellations were computer-simulated with fading coefficients that were chosen randomly but held constant for two consecutive matrix-valued signals. In one exceptional case described below, the constellation was transmitted over a functional three-transmitter-antenna wireless channel. The resulting figures plot the block probability of decoding a matrix incorrectly, denoted P e . FIG. 2 displays the simulated performance of the group SL 2 ( F 5 ) which has 120 elements, and therefore has a rate R = log(120)/2≈3.45. We also compare the best Abelian group we could find, a two-dimensional orthogonal design as described in V. Tarokh et al., "A differential detection scheme for transmit diversity," to be published in J. Sel. Area Comm . (2000), and the quaternion group Q 6 , having L = 128 matrices and rate R = 3.5. The orthogonal design had 121 elements and was parametrized by a 2 × 2 matrix filled with 11 th roots of unity. The excellent performance of SL 2 ( F 5 ) is evidenced by the aproximately 2.5 dB improvement over the orthogonal design (which is not a group), the 6.5 dB improvement over the Abelian group, and the 13 dB improvement over the quaternion group. FIG. 3 is the same as FIG. 2 except that we now assume N = 2 receive antennas. The relative merits of the various constellations are now more pronounced. The advantage in performance of the N = 2 system compared to the N = 1 system is also apparent. FIG. 4 provides a similar comparison to FIG. 2 except that we are now considering rate R ≈ 4. The group constellation is F 15,1,11 with L = 240 elements ( R = 3.95). The other constellations are the best orthogonal design, diagonal constellation and quaternion groups of comparable rate. FIG. 5 shows the performance advantage of the M = 3 antenna, 63-element (R = 1.99) group G 21,4 compared to the best three transmit antenna 63-element diagonal constellation. We were also able to transmit this constellation over a wireless apparatus located within a hallway at our research facility. The three transmit antennas were separated from the one receive antenna by approximately 10 meters around a bend in the hallway lined with metal walls and equipment, thus creating a rich quasi-static scattering environment. FIG. 6 shows the performance. FIG. 7 shows the advantage of K 1,1,-1 for M = 4 transmitter antennas, and compares it with the best Abelian group we found. Again, the performance advantage of this group over the Abelian group is evident. Table 2 collects together some of the group constellations that we have found with high ζ for different numbers of antennas M and rates R . M L R ζ Comments - 2 - 1 { I, -I } for any M 2 24 2.29 0.5000 E 3,1 = SL 2 (F 3 ) 2 48 2.79 0.3868 F 3,1,-1 2 120 3.45 0.3090 J 1,1 = SL 2 (F 5 ) 2 240 3.95 0.2257 F 15,1,11 3 9 1.06 0.6004 Cyclic group G 9,1 with u = (1,2,5) 3 63 1.99 0.3851 G 21,4 3 513 3.00 0.1353 G 171,64 ( t = 19) 3 4095 4.00 0.0361 G 1365,16 ( t = 91) 3 32445 5.00 0.0131 G 10815,46 ( t = 721) 4 240 1.98 0.5000 K 1,1,-1 5 1025 2.00 0.1679 G 205,16 ( t = 41) 5 33825 3.01 0.0501 G 6765,16 ( t = 451) 5 1021025 3.99 0.0037 G 204205,21 ( t = 40841) 7 16513 2.00 0.0955 G 2359,8 ( t = 337) 9 513 1.00 0.361 G 57.4 Reducible Representations It should be noted that in at least some cases, signal constellations may be based upon reducible, rather than irreducible, group representations. In particular, given an irreducible representation of a group, consisting of matrices A 1 ,..., A L , a reducible representation may be derived therefrom by block diagonal construction. For example, given an invertible, complex linear transformation T of appropriate dimension, a corresponding reducible representation consists of the matrices A more general construction for deriving a reducible representation of the group is defined in terms of some number m of automorphisms  i of the group, i = 1,..., m . An automorphism is a one-to-one mapping of the group onto itself that preserves the identity element and respects multiplication; i.e., the image of a product of two multiplicative factors under the mapping is the product of the images of the respective factors. According to such a construction, each of the matrices A i is replaced by an m×m matrix of the form: A construction for deriving a further constellation from any of those described above is defined in terms of arbitrary, but fixed, unitary matrices U and V . Given the constellation of matrices the new constellation consists of the matrices It should be noted that the new constellation does not necessarily form a group. Herein, we define the term coset to include any set derived from a group or from a subset of a group by left multiplication, right multiplication, or left and right multiplication. Thus, according to our definition, the constellation consisting of the matrices is a coset of the constellation of matrices

Claims: A method of constructing a space-time constellation, which consists of a set of square matrices of common size at least 2 × 2, and which has a positive diversity product; where the diversity product is defined as the minimum value of (½) det (V - V') | 1/N , with N equal to the number of rows in each of the matrices, as (V, V') ranges over all distinct pairs of said matrices; CHARACTERISED BY selecting a fixed-point free non-Abelian group, other than a quaternion group; selecting a fixed-point free matrix representation of said group; taking a non-empty subset of said representation, thereby producing a set M of matrices; multiplying all members of M by a common scalar, thereby producing a set of matrices M'; right multiplying all the members of M' by a common matrix, thereby producing a set of matrices M"; and left multiplying all the members of M" by a common matrix, thereby producing the space-time constellation. A method as claimed in claim 1, wherein: given a pair of integers (m, r ), n is the smallest positive integer greater than 1 such that, r n ≡ 1 mod m, r 0 is defined to be the greatest common divisor of r -1 and m.t is defined to be m / r 0 , and the pair (m, r ) is said to be admissible if the greatest common divisor of n and t is 1; and the non-Abelian group belongs to the class G m.r , ( m, r ) is admissible, the number of elements in the group is mn , and the number of antennas used for transmission is n . A method as claimed in claim 1, wherein: given a pair of integers ( m , r ), n is the smallest positive integer greater than 1 such that, r n ≡ 1 mod m , r 0 is defined to be the greatest common divisor of r -1 and m, l is defined to be m / r 0 , and the pair (m, r) is said to be admissible if the greatest common divisor of n and t is 1; nr 0 is even, s is the highest power of 2 that divides mn , and at least one integer I satisfies l 2 ≡ 1 mod m, l ≡ 1 mod n, l ≡ -1 mod s ; and the non-Abelian group belongs to the class D m.r.l , ( m, r ) is admissible, the number of elements in the group is 2mn, and the number of antennas used for transmission is 2 n . A method as claimed in claim 1, wherein: given a pair of integers ( m , r ), n is the smallest positive integer greater than 1 such that, r n ≡ 1 mod m, r 0 is defined to be the greatest common divisor of r - 1 and m , t is defined to be m / r 0 , and the pair (m, r ) is said to be admissible if the greatest common divisor of n and t is 1; mn is odd and nr 0 is divisible by 3; and the non-Abelian group belongs to the class E m , r , ( m , r ) is admissible, the number of elements in the group is 8 mr , and the number of antennas used for transmission is 2 n . A method as claimed in claim 1, wherein: given a pair of integers ( m.r ), n is the smallest positive integer greater than 1, such that, r n ≡ 1 mod m , r 0 is defined to be the greatest common divisor of r -1 and m , t is defined to be m / r 0 , and the pair ( m , r ) is said to be admissible if the greatest common divisor of n and t is 1; mn is odd, r 0 is divisible by 3, n is not divisible by 3, and at least one integer l satisfies: l 2 ≡ 1 mod m, l ≡ 1 mod n, l ≡ -1 mod 3; and the non-Abelian group belongs to the class F m , r.l , (m, r ) is admissible, the number of elements in the group is 16 mn , and the number of antennas used for transmission is 4 n . A method as claimed in claim 1, wherein: given a pair of integers ( m , r ), n is the smallest positive integer greater than 1 such that, r n ≡ 1 mod m, r 0 is defined to be the greatest common divisor of r -1 and m, t is defined to be m / r 0 , and the pair ( m , r ) is said to be admissible if the greatest common divisor of n and t is 1; the greatest common divisor of mn and 120 is 1; and the non-Abelian group belongs to the class J m.r , ( m , r ) is admissible, the number of elements in the group is 120 mn , and the number of antennas used for transmission is 2 n. A method as claimed in claim 1, wherein: given a pair of integers ( m , r ), n is the smallest positive integer greater than 1 such that r n ≡ 1 mod m ; at least one integer l satisfies: l 2 ≡ 1 mod m , l ≡ 1 mod n ; and the non-Abelian group belongs to the class K m.r.l , the number of elements in the group is 240mn, and the number of antennas used for transmission is 4 n .

Inventors: HASSIBI BABAK, HOCHWALD BERTRAND M, SHOKROLLAHI MOHAMMAD AMIN, SWELDENS WIM, HASSIBI, BABAK, HOCHWALD, BERTRAND M., SHOKROLLAHI, MOHAMMAD AMIN, SWELDENS, WIM
Applicants: LUCENT TECHNOLOGIES INC, LUCENT TECHNOLOGIES INC.
Application year: 2000
Main classification: H04L1
IPC codes: H04B7, H04L1