TRANSMISSION OVER BUNDLED CHANNELS IN A CDMA MOBILE RADIO SYSTEM

Abstract: A transmitter apparatus and transmitting method are provided for implementing a high data rate multi-channel link in a digital Code Division Multiple Access (CDMA) wireless communications system composed of three or more individual lower data rate traffic channels represented in a transmitted combined RF signal, which are distributed over a set of phase offsets of a sinusoid carrier in the range of 0 to pi radians, in which a combined complex spread spectrum signal, composed of an in-phase component and a quadrature-phase component, is formed prior to up-conversion in a manner that all the required phase offsets of the represented channels are introduced. The three or more individual lower data rate channels include two or more individual channels having phase offsets which are 0, pi /2, or pi radians, and one or more individual channels having phase offsets which are pi /4 or 3 pi /4 radians. Individual complex spread spectrum signals are formed for the respective individual channels, each composed of an in-phase component and a quadrature-phase component. Further complex spread spectrum signals are derived therefrom (or merely repeated) for the individual channels in a manner including the introduction of fractional scaling factors having an absolute value substantially equal to √ 2ROOT 2/2 for the channels having phase offsets of (/4 or 3(/4 radians, and these derived signals are additively combined to form the complex combined spread spectrum signal.

Description: Transmission over bundled channels in a CDMA mobile radio system. BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to apparatus for, and methods of, carrying data for a same communication over a plurality or bundle of traffic channels of a communications system in order to achieve a data rate or bandwidth for the communication which is a multiple of that provided by a single traffic channel. In its particular aspects, the present invention relates to the transmitter architecture of a multi-channel reverse link in a digital Code Division Multiple Access (CDMA) wireless communications system, wherein individual channels represented in a transmitted combined RF signal are distributed over a set of phase offsets in order to reduce the peak-to-average ratio of the combined RF signal over that which would occur if the carrier phases of the individual channels were aligned. 2. Description of the Related Art Such a system is know from: a submission by E. Tiedemann et al. of Qualcomm Incorporated entitled "Improving the Reverse Link Peak To Average Ratio (Phase lc)" to the TR 45.5 subcommittee of the Telecommunications Industry Association (TIA) at a meeting held in Philadelphia, PA, June 16-20, 1997; from TIA/EIA interim standard IS- 95B; and also from PCT patent application WO 98/58457. In 1992, a direct sequence code division multiple access (DS-CDMA) system was adopted as interim standard IS-95 by the TIA in association with the Electronics Industry Association (EIA) for deployment in the cellular band at 800MHz. After successful field tests and trial systems the IS-95 system is now operating with tens of millions of subscribers. CDMA is based on spread spectrum technology originally developed by the Allies during World War II to resist enemy radio jamming. Spread spectrum signals are characterized by a bandwidth W occupied by signals which is much greater than the information rate R of the signals in bit/s. Thus, a spread spectrum signal inherently contains a kind of redundancy which can be exploited for overcoming several kinds of interference (including signals from other users in the same band and self-interference in the sense of delayed multipath components). Another key property of spread spectrum signals is pseudo-randomness. Therefore, the signal appears to be similar to random noise, making it difficult to demodulate by receivers other than the intended ones. In CDMA systems, channels share a common bandwidth and are distinguished by different code sequences. In the case of IS-95, each channel carries an encoded information signal which has modulated by a specific one of a set of orthogonal sequences, known as Walsh codes, which is assigned to the channel (known as applying a Walsh cover), further modulated or scrambled by long Pseudo Noise (PN) codes, modulated by in-phase and quadrature-phase PN short codes to form respective in-phase (I) and quadrature-phase (Q) spread spectrum signal components of a complex spread spectrum signal for direct up-conversion by multiplication of these in-phase and quadrature-phase components with in-phase and quadrature-phase sinusoids at the carrier frequency, respectively, addition of the in-phase and quadrature-phase results of the multiplication to form an RF signal, which is then amplified in an RF power amplifier and supplied to an antenna. IS-95 has been extended to interim standard IS-95 A in 1995, and more recently to interim standard IS-95B. The last extension provides for high bandwidth data applications where a set or bundle of up to eight channels can be used to carry data from the same communication, in effect forming a high data rate channel from the set of lower data rate channels. IS-95B provides that when a plurality of channels are used to form a multi- channel link, that in the reverse link (transmission from mobile station to base station) the pairs of in-phase and quadrature-phase sinusoids at the carrier frequency for these channels are distributed in phase offset over the range of 0 to π radians in a particular sub-optimal manner in order to reduce the peak-to-average power of the combined RF signal over what be the case if the pairs of sinusoids had the same zero phase offset for each channel. This distribution of phase offset reduces the linearity and dynamic range requirements of the power amplifier in the mobile station. Specifically, the phase offsets of the in-phase and quadrature-phase sinusoids applied to up-convert channels 0-3 and also channels 4-7 are 0, π/2, π/4, 3π/4 radians, respectively. As is well known, the in-phase and quadrature-phase sinusoids at the carrier frequency are analog signals produced by an oscillator, which are analog multiplied with the in-phase and quadrature-phase spread-spectrum signal components of the channels after the latter are converted from digital to analog signals. The prospect of using a separate D/A converter in, or providing for separate D/A conversion for, the in-phase and quadrature- phase spread spectrum signal components of each of up to eight channels adds complexity and cost to the mobile station. Further, while in-phase and quadrature-phase sinusoids are generated from the oscillator section of the mobile station, the stable generation of further sinusoids with phase offsets of π/4 and 3π/4 is problematic. The aforementioned PCT patent application shows that where only two channels are used in combination, the π/2 phase offset of the second channel may be introduced by summing the in-phase spread spectrum signal component of the first channel and the negative of the quadrature-phase spread spectrum signal component of the second channel to form a combined in-phase spread spectrum signal component, and summing the quadrature-phase spread spectrum signal component of the first channel and the in-phase spread spectrum signal component of the second channel to form a combined quadrature- phase components. The combined in-phase and combined quadrature-phase components are then directly up-converted by multiplication with in-phase and quadrature-phase sinusoids at the carrier frequency, respectively, and these products are added to form the combined RF signal. However, where three or more channels are used in combination, necessitating the introduction of a π/4 phase offset, and possibly also a 3π/4 phase offset, a set of in-phase and quadrature-phase sinusoids is used, the set including at least in-phase and quadrature-phase sinusoids having a π/4 phase offset. OBJECTS AND SUMMARY OF THE INVENTION It is an object of the present invention to provide an architecture for a multichannel link in a digital Code Division Multiple Access (CDMA) wireless communications system, of a type wherein a plurality of individual traffic channels represented in a transmitted combined RF signal are distributed over a set of phase offsets of the sinusoid carrier in the range of 0 to π radians, which requires only D/A conversion of a combined in- phase spread spectrum signal component and a combined quadrature-phase spread spectrum signal component of a combined complex spread spectrum signal, and the generation of only one pair of in-phase and quadrature carrier-phase sinusoids at the carrier frequency for multiplication with different ones of a pair of signals derived from the combined in-phase and combined quadrature-phase spread-spectrum signal components. Such an architecture would simplify and reduce the cost of baseband and RF sections in such a mobile station capable of using a multi-channel link composed of three or more individual channels. This and other objects of the present invention are satisfied by providing a transmitter apparatus and transmitting method in which all the required phase offsets, including those of two or more of the represented individual channels having values which are 0, π/2, or π and those of one or more of the represented individual channels having values which are not 0, π/2, or π radians, are introduced in the formation of combined in-phase and combined quadrature-phase spread spectrum signal components. To accomplish this, individual complex spread spectrum signals for the respective individual channels, each composed of in-phase component and a quadrature-phase component, are first formed, and signals derived from the individual complex spread spectrum signals for the respective individual channels are additively combined to form the combined complex spread spectrum signals. This additive combination is such that signals derived from the individual complex spread spectrum signals for the one or more channels having phase offsets which are not 0, π/2, or π radians are derived by applying fractional scaling factors relative to the scale of the signals derived from the individual complex spread spectrum signals for the two or more channels having phase offsets which are 0, π/2, or π. More specifically, those values of phase offset which are not 0, π/2, or π are π/4 or 3 π/4, and the fractional scale factors have an absolute value of substantially V2 /2. Further, the means for, or acts of, additively combining are arranged such that the in-phase and quadrature phase components, respectively, of the combined spread spectrum signal receive contributions from the signals derived from either but not both of the in-phase or quadrature phase components of the complex individual spread spectrum for the channels having phase offsets which are 0, π/2, or π, while they receive contributions from signals derived from both of the in-phase or quadrature phase components of the complex individual spread spectrum for the channels having phase offsets which are not 0, π/2, or π. In accordance with a first embodiment of the invention, the in-phase and quadrature-phase components of each of the complex spread spectrum signals for the respective individual channels are applied to the inputs of respective finite impulse response (FIR) filters or filter operations, the outputs or results of which feed the means for, or acts of, additively combining. In accordance with a second embodiment of the invention, the in-phase and quadrature-phase components of each of the complex spread spectrum signals for the respective individual channels are applied to the inputs of respective finite impulse response (FIR) filters or filter operations, the outputs or results of which feed the means for or acts of additively combining. This embodiment has the advantage of requiring only two FIR filters or filtering operations, rather than two per individual channel. Other objects, features and advantages of the present invention will become apparent upon perusal of the following detailed description when taken in conjunction with the appended drawing, wherein: BRIEF DESCRIPTION OF THE DRAWING Figure 1 is a general schematic of a mobile station or handset for use in a digital cellular system, e.g. a CDMA system; and Figures 2 and 3 are functional schematics of a portion of a transmitter embodied in the mobile station of Figure 1 in accordance with the first and second embodiments of the invention, respectively. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Referring first to Figure 1 of the drawing, a mobile station transceiver 10 for a digital cellular system, e.g. a CDMA system, which at the level of detail shown is conventional, comprises a user interface section 12 coupled to a baseband section 14, which is coupled to an RF section 16. User interface section 12 is also coupled to a keypad 18, LCD display 20, microphone 22, and speaker 24. Baseband section 14 includes a digital signal processor (DSP) 26, which has access to a random access memory and to a read only memory 30 containing firmware instructions. Also included in baseband section 14 are a clock 32, analog-to-digital converters (A/D's) 34 notably for receiving analog signals from RF section 16 in the forward (reception) link and digital-to-analog converters (D/A's) 34 notably for supplying analog signals to RF section 16 in the reverse (transmission) link. RF Section 16 comprises an oscillator 38 for deriving sinusoids at the carrier frequency and supplying them to zero IF frequency down-converter 40 in the forward link and zero IF frequency up-converter 42 in the reverse link. Down-converter 40 receives an RF signal from an input amplifier 44 which is fed from an antenna 48 via a diplexer 46, whereas up-converter 42 supplies an RF signal to an output or power amplifier 50, which feeds antenna 48 via diplexer 46. As will soon appear from the description of the first and second embodiments of the invention, all of the required carrier phase offsets for forming a multi-channel reverse link composed of three or more channels are implemented in the processing by baseband section 14 in DSP 32, enabling the up-converter 42 in RF section 16 to comprise a pair of mixers or multipliers for multiplying in-phase and quadrature-phase components of a complex combined signal with in-phase and quadrature-phase sinusoids at the carrier frequency. Turning to Figure 2 in regard to the first embodiment, there is shown a block 52, implemented by baseband section 14, which receives data signals DSo - DSN for the respective N channels forming the multi-channel reverse link, and forms an analog complex combined spread spectrum signal consisting of an analog combined spread spectrum in-phase component ACSSi and an analog combined spread spectrum quadrature-phase component ACSSQ. While under IS-95B up to eight channels can form a multi-channel link, only four are shown for ease of illustration. The arrangement of the baseband processing for channels 0-3 is the same as for channels 4-7, since IS-95B provides that the phase offsets of the in- phase and quadrature-phase sinusoids applied to up-convert channels 0-3 and channels 4-7 follow the same sequence, namely 0, π/2, π/4, 3π/4 radians. It will be evident that such additional channels merely add further contributions into the formation of the analog combined spread spectrum signal component pair ACSSi and ACSS Q . The latter components feed zero IF up-converter 54 implemented in RF section 16. In up-converter 54, the analog combined in-phase and quadrature-phase spread spectrum signal components, ACSSi and ACSSQ, are applied to low pass filters 140 and 141, respectively, and the outputs thereof are applied to mixers or multipliers 144 and 146, respectively, as are in-phase and quadrature- phase sinusoids, sin(ωct) and cos(coct), at the carrier frequency. The outputs of mixers or multipliers 144 and 146 are applied to an adder 150 in order to form a combined RF signal CRF for supply to power amplifier 50. Baseband processing block 52 comprises a block 58 which forms digital individual in-phase and quadrature-phase spread spectrum signal component pairs DSSoi, DSSOQ, through DSS 3 ι, DSS 3Q in a conventional manner in response to the individual input data signals DS 0 to DS 3 , respectively, and a block 56 which filters, scales, and combines the digital individual in-phase and quadrature-phase spread spectrum signal component pairs DISSoi, DISS O Q, through DISS ι, DISS Q to form the digital combined in-phase and quadrature-phase spread spectrum signal component pair DCSSi and DCSSQ in a manner which introduces or accounts for the effect of the required carrier phase offsets of the individual channels. The digital combined components DCSSi and DCSSQ are applied to digital-to-analog converters 140 and 142, respectively, to form the analog combined components, ACSSj and ACSS Q . In block 58 the individual input data signals DS 0 , DSi, DS 2 , and DS 3 are applied to respective channel encoders 61, 71, 81, and 91, and the resultant frame-wise encoded data signals are applied to respective Walsh modulators 62, 72, 82, and 92 to modulate the encoded data signals with respective Walsh codes specific to the individual channels. The resultant Walsh code modulated encoded data signals are then applied to respective multipliers (or modulo-2 adders) 63, 73, 83, and 93 where they are modulated by respective pseudo noise (PN) sub-sequences which may have different starting positions in a continually generated PN long code sequence typically having a period of 2 42 -l to spread the spectra of the Walsh code modulated encoded signals. Next, the long code modulated signals are applied to respective multipliers or modulo-2 adders 64, 74, 84, and 94 where they are modulated by an in-phase PN short code sequence, PN_I to produce the respective spread spectrum in-phase signals DSSoi, DDSn, DSS ι, and DSS 3 ι, and also to respective multipliers or modulo-2 adders 65, 75, 85, and 95 where they are modulated by a quadrature-phase PN short code sequence, PN_Q. The results of modulation with the quadrature-phase PN short code sequence PN_Q are applied to respective 1/2 chip delays 66, 76, 86, and 96 to produce the respective spread spectrum quadrature-phase signals DSS OQ , DDS IQ , DSS Q, and DSS 3 Q. The in-phase and quadrature-phase short code sequences PN_I and PN_Q typically have a period of 2 15 -1 and are specific to the base station which the mobile station is communicating. Also, in block 58, a data burst randomizer 69b is fed by the PN long code subsequence applied in channel 0, which in turns produces a control signal after a delay 70 to turn on power amplifier 50 as needed. Block 56 includes a block 56a which derives signals DDSSoi, DDSSπ, DDSS 2 ι, and DDSS 3 ι from signals DSS 0 ι, DSS π , DSS 2 ι, and DSS 3 ι by application of the latter to respective finite impulse response shaping filters (FIR_I) 67, 77, 87, and 97, and also derives signals DDSS 0Q , DDSS JQ , DDSS 2Q , and DDSS 3Q from signals DSS 0Q , DSS I Q, DSS 2Q , and DSS 3Q by application of the latter to respective finite impulse response shaping filters (FIR_Q) 668, 78, 88, and 98. It should be noted that FIR_I 77 introduces a scaling factor of -1, FIR_I 87, FIR_Q 88, and FIR_Q 98 introduce a fractional scaling factor of substantially V2 /2, and FIR_I 97 introduces a fractional scaling factor of substantially -V2 /2. Block 56 further includes a block 56b which additively combines the derived signals DDSS 0 t, DDSS 0Q , DDSS U , DDSS 1Q , DDSS 2 ι, DDSS 2Q , DDSS 3I , and DDSS 3Q to form the digital combined signal components DCSSi and DCSS Q . The term "additively combining" is intended to include subtraction and/or changing the sign of an operand prior to addition. Block 56b is illustrated as comprising an adder 89 which forms an intermediate spread spectrum in-phase signal for channel 2, IDSS 2 ι, by adding derived signals DDSS 2 I and DDSS 2Q and an adder 90 which forms an intermediate spread spectrum quadrature-phase signal for channel 2, IDSS 2Q , by subtracting derived spread spectrum DDSS 2 ι from derived spread spectrum signal DDSS 2Q . Block 56b further comprises an adder 99 which forms an intermediate spread spectrum in-phase signal for channel 3, IDSS 3 ι, by adding derived signals DDSS 3 ι and DDSS 3Q and an adder 100 which forms an intermediate spread spectrum quadrature-phase signal for channel 3, IDSS 3Q , by subtracting derived spread spectrum DDSS Q from derived spread spectrum signal DDSS ι. It should be appreciated that the combination of the fractional scaling factors introduced in block 56a having an absolute value of substantially V212 together with the sums and differences formed by the adders 89, 90, 99, and 100 introduce into the intermediate signal pair IDSS 2 ι, IDSS 2Q the effect of a phase rotation by π/4 radians, introduce into the signal pair IDSS 3 ι, IDSS 3Q the effect of a phase rotation by 3π/4 radians. Further, with regard to channel 1, the effect of a phase rotation by π/2 is introduced merely by swapping the in-phase and quadrature phase components, since a needed sign change of the in-phase component prior to the swap has been incorporated in the -1 scaling factor of FIR I 77. The in-phase and quadrature-phase contributions from the various channels, after accounting for all the required phase offsets, are summed in respective adders 110 and 120. whose outputs are the respective combined in-phase and quadrature-phase signal components, DCSSi and DCSSQ. Since pursuant to IS-95B it is possible to bundle up to eight channels, actually adders 1 10 and 120 have additional inputs (not shown) for channels 4-7, which as previously mentioned are the same form as for channels 0-4, respectively. As illustrated, adder 110 receives as inputs the signals DDSSoi, DDSSIQ, IDSS 2 I, and IDSS 2Q , whereas adder 120 receives as inputs the signals DDSS 0Q , DDSSn, IDSS 2 Q, and IDSS 3 Q. Adders 110 and 120 also receive a same scale control signal on respective lines 115, 125 to maintain the proper dynamic range for the result signals DCSSi and DCSSQ. It should be appreciated that the blocks 56 could take a variety of forms. The required fractional scaling factors can be introduced at different locations therein (for example at the pertinent inputs to adders 110 and 120), and a scaling factor of-1 can be introduced as a sign change prior to addition. Further, the various additions and subtractions could be changed in their order, further broken down, or further combined. For example, adders 89 and 99 could be eliminated with their functions being incorporated into adder 110 and adders 99 and 100 could be eliminated with their functions being incorporated into adder 120. In such a case adders 110 and 120 would each have six inputs receiving one contribution from each of channels 0 and 1 and receiving two contributions from each of channels 2 and 3. Preferably, the digital data for all operations in block 58 including the inputs to the FIR filters of block 56a are single-bit bipolar, taking the values +1, whereas the outputs of the FIR filters of block 56a. and all subsequent digital data up to and including the outputs of adders 1 10 and 120 are five-bit bipolar taking the integer values in the range of -15 to +15. Figure 3 shows a second embodiment which appears the same as the embodiment of Figure 2 except that block 56 in Figure 2 comprised of blocks 56a and 56b, is replaced by a block 156 in Figure 3 comprised of block 156a (which does not contain FIR filters and merely introduces the required fractional scale factors), a block 156b (in which adder 120 changes the sign of DDSSn prior to addition to introduce the needed -1 factor), and added shaping filters FIR_I 167 and FIR_Q 168 which receive the five bit bipolar outputs of the respective adders 110, 120 and form the respective five bit bipolar combined signals DCSSi and DCSS Q . The consequent replacement of the digital FIR filters 67, 68, 77, 78, 87. 88, 97, 98 of Figure 2 in the in-phase and quadrature-phase paths of the individual channels with the digital filters 167 and 168 of Figure 3 in the in-phase and quadrature-phase paths of the combined signals further reduces complexity, notwithstanding that the filters 167 and 168 take five-bit inputs. It should now be appreciated that the objects of the present invention have been satisfied. While the present invention has been described in particular detail, it should also be appreciated that numerous modifications are possible within the intended spirit and scope of the invention. In interpreting the appended claims it should be understood that: a) the word "comprising" does not exclude the presence of other elements or steps than those listed in a claim; b) the word "a" or "an" preceding an element does not exclude the presence of a plurality of such elements. c) any reference signs in the claims do not limit their scope; and d) several "means" may be represented by the same item of hardware or software implemented structure or function.

Claims: CLAIMS: 1. A transmitter apparatus for a high data rate multi-channel link in a digital Code Division Multiple Access (CDMA) wireless communications system, wherein a plurality of individual lower data rate traffic channels represented in a transmitted combined RF signal (CRF) are distributed over a set of phase offsets of a sinusoid carrier in the range of 0 to π radians, comprising: means for forming a combined complex spread spectrum signal composed of an in-phase component (DCSSI or ACSSI) and a quadrature-phase component (DCSSQ or ACSSQ), in a manner that all the required phase offsets of the represented channels are introduced; and means (144, 146, 150) for up-converting the combined complex spread spectrum signal by multiplication (144) of a signal derived from its in-phase component (DCSSQ or ACSSI) with an in-phase sinusoid (sin(act)) and multiplication (DCSSQ or ACSSQ) of a signal derived from its quadrature-phase component (DCSSQ or ACSSQ) with a quadrature-phase sinusoid (cos(act)), and addition (150) of the results of the multiplications ( 144, 146) to form the combined RF signal (CRF); wherein said plurality of individual lower data rate channels are three or more individual channels (channels 0, 1, 2, 3), including two or more individual channels (channels 0, 1) having phase offsets which are 0, π/2, or π radians, and one or more individual channels (channels 2, 3) having phase offsets which are not 0, π/2, or π radians. 2. The apparatus as claimed in Claim 1, wherein the means for forming a combined complex spread spectrum signal comprises: means (58) for forming individual complex spread spectrum signals for the respective individual channels, each composed of in-phase component (DSS0I, DSS1I, DSS2I, DSS3I) and a quadrature-phase component DSSOQ, DSSIQ , DSS2Q, DSS3Q); and means (56b, 156b) for additively combining signals derived (56a, 156a) from the individual complex spread spectrum signals for the respective individual channels to form the combined complex spread spectrum signals; wherein the signals derived from the individual complex spread spectrum signals for the one or more channels (channels 2, 3) having phase offsets which are not 0, π/2, or π radians are derived (56a, 15όa) by applying fractional scaling factors relative to the scale of the signals derived from the individual complex spread spectrum signals for the two or more channels (channels 0, 1) having phase offsets which are 0, π/2, or π. 3. The apparatus as claimed in Claim 2, wherein the means (56b) for additively combining is configured such that the in-phase (DCSSI or ACSSI) and quadrature phase components (DCSSQ or ACSSQ), respectively, of the combined spread spectrum signal receive contributions from the signals derived from either but not both of the in-phase or quadrature phase components of the complex individual spread spectrum for the channels (channels 0, 1) having phase offsets which are 0, π/2, or π. while they receive contributions from signals derived from both of the in-phase or quadrature phase components of the complex individual spread spectrum for the channels (channels 2,3) having phase offsets which are not 0, π/2, or π. 4. The apparatus as claimed in Claim 2, wherein said phase offsets which are not 0, π/2, or π are π/4 or 3π/4, and the fractional scale factors have an absolute value of substantially ^ 2 /2. 5. The apparatus as claimed in Claim 3, wherein said phase offsets which are not 0, π/2. or π are π/4 or 3π/4, and the fractional scale factors have an absolute value of substantially ^ 2 I2. 6. The apparatus as claimed in Claims 1-5, wherein the in-phase (DSS0I, DSS1I, DSS2I, DSS3I) and quadrature-phase components (DSSOQ, DSSIQ, DSS2Q, DSS3Q) of each of the complex spread spectrum signals for the respective individual channels are applied to the inputs of respective FIR filters (67, 68, 77, 78, 87, 88, 97. 98) the outputs of which feed the means (56b) for additively combining. 7. The apparatus as claimed in Claims 1-5, wherein the in-phase (DCSSI) and quadrature-phase (DCSSQ) components of the combined complex spread spectrum signal are applied to the inputs of respective FIR filters (167, 168), the outputs of which feed the means (144, 146. 150) for up-converting. 8. A transmitting method for a high data rate multi-channel link in a digital Code Division Multiple Access (CDMA) wireless communications system, wherein a plurality of individual lower data rate traffic channels represented in a transmitted combined RF signal (CRF) are distributed over a set of phase offsets of a sinusoid carrier in the range of 0 to π radians, comprising: forming a combined complex spread spectrum signal composed of an in-phase component (DCSSI or ACSSI) and a quadrature-phase component (DCSSQ or ACSSQ), in a manner that the phase offsets of each of the represented channels are introduced; and up-converting (144. 146. 150) the combined complex spread spectrum signal by multiplication (144) of a signal derived from its in-phase component (DCSSI or ACSSI) with an in-phase sinusoid (sin(act)) and multiplication (146) of a signal derived from its quadrature-phase component (ACSSI or ACSSQ) with a quadrature-phase sinusoid (cos(act)), and addition (150) of the results of the multiplications (144, 146) to form the combined RF signal (CFR); wherein said plurality of individual lower data rate channels are three or more individual channels, (channels 0, 1, 2, 3) including two or more individual channels (channels 0, 1) having phase offsets which are 0, π/2, or π radians, and one or more individual channels (channels 2, 3) having phase offsets which are not 0, π/2, or π radians. 9. The method as claimed in Claim 8, wherein said act of forming a combined complex spread spectrum signal comprises: forming individual complex spread spectrum signals for the respective individual channels, each composed of in-phase component (DSS0I, DSS1I, DSS2I, DSS3I) and a quadrature-phase component (DSSOQ, DSSIQ, DSS2Q, DSS3Q); and additively combining (56b, 156b) signals derived from the individual complex spread spectrum signals for the respective individual channels to form the combined complex spread spectrum signals; wherein the signals derived from the individual complex spread spectrum signals for the one or more channels (channels 2, 3) having phase offsets which are not 0, π/2, or π radians are derived by applying fractional scaling factors relative to the scale of the signals derived from the individual complex spread spectrum signals for the two or more channels (channels 0, 1) having phase offsets which are 0, π/2, or π. 10. The method as claimed in Claim 2, wherein said act of additively combining (56b, 156b) is such that the in-phase and quadrature phase components, respectively, of the combined spread spectrum signal receive contributions from the signals derived from either but not both of the in-phase or quadrature phase components of the complex individual spread spectrum for the channels (channels 0, 1) having phase offsets which are 0, π/2, or π, while they receive contributions from signals derived from both of the in-phase or quadrature phase components of the complex individual spread spectrum for the channels (channels 2, 3) having phase offsets which are not 0, π/2, or π.

Inventors: LI XIAO B, LI, XIAO, B.
Applicants: KONINKL PHILIPS ELECTRONICS NV, KONINKLIJKE PHILIPS ELECTRONICS N.V.
Application year: 2000
Main classification: H04B1
IPC codes: H04B1, H04Q7