Pulse amplifier with low duty cycle errors

Abstract: A low power pulse amplifier with low duty cycle errors. The amplifier provides several differential amplifier stages with a biasing and canceling network. To minimize duty cycle errors for large input signals, cascode transistors are added between the drains of the differential amplifiers and the outputs. The result is an amplifier having a duty cycle error of less than 5% at amplitude input ranges from 5 millivolts to the supply voltage.

Description: The present invention relates to a low power pulse amplifier, and more particularly to a low power pulse amplifier with a low duty cycle error at high input levels. The duty cycle requirements for pulse amplifiers of this type exclude the use of single-ended amplifiers since clipping at higher amplitudes results in different clip levels and delays in the positive and negative directions. Typically, for a single-ended input signal, an ordinary differential stage consisting of a differential pair of transistors produces asymmetrical delays in the positive and negative directions of the input signal if the input signal is large. The main reason for this is that the common source node for the differential pair will change its potential differently between large positive and large negative input signals. When the input signal turns off the input transistor, the voltage of the common source node only changes a small amount due to the doubled current in the other transistor. In the other direction the common source node shorts to the output node and changes a substantial amount. One solution to this problem is to reduce the absolute delay times by increasing bandwidth. But this solution comes at the expense of increased current consumption. GB-A-2,258,779 and EP-A-0,451,378 both disclose a pulse amplifier with a plurality of fully differential amplifier stages between an input and output. GB-A-2,258,779 employs complex circuitry that is not suited for low power applications. EP-A-0,451,378 addresses the problem of negative overshoot caused by capacitance at the common node and employs a bias voltage applied to the drain of the differential transistors. An object of the present invention is to provide an improved pulse amplifier. According to the present invention there is provided a pulse amplifier comprising an input for receiving input signals, an output, and a plurality of cascaded fully differential amplifier stages between said input and output, wherein each said amplifier stage comprises first and second transistors forming a differential pair and having a common node, and a pair of cascode transistors in series with said respective first and second transistors, each of said transistors having first and second main electrodes and a gate electrode, the gate electrodes of said first and second transistors receiving input signals for the amplifier stage, the second main electrodes of said first and second transistors being connected to the first main electrodes of said respective cascode transistors, characterized in that said amplifier further comprises a first series combination comprising a first load transistor in series with a first current source and having a first common connection point, and a second series combination comprising a second load transistor in series with a second current source and having a second common connection point, said first and second common connection points providing output signals for said amplifier stage and being connected to the second main electrodes of said respective cascode transistors, and wherein in the presence of large input signals one of the transistors of said differential pair is driven hard into conduction causing the first main electrode of its associated cascode transistor to be substantially shorted to said common node, thereby causing the voltage of the first main electrode of the associated cascode transistor to approach its gate voltage, and in turn causing a gradual reduction in the current flow through said associated cascode transistor and consequently a gradual reduction in the current flow through said one of the transistors of the differential pair that was driven into hard conduction. In accordance with the invention fully differential gain stages, that is differential amplifiers with differential inputs and differential outputs, give more symmetrical delays and reduce the problem. In one embodiment of the invention, the first and second current sources are connected in series with the series combinations of the first and second transistors and their associated cascode transistors, and the first and second load transistors are connected in parallel with said series combinations of said first and second transistors and their associated cascode transistors. In an alternative embodiment of the invention the first and second current sources are connected in parallel with said series combinations of the first and second transistors and their associated cascode transistors, and the first and second load transistors are connected in series with said series combinations of first and second transistors and their associated cascode transistors. The invention also preferably includes a biasing network and an offset canceling network. The invention will now be described in more detail, by way of example only, with reference to the accompanying drawings, in which:- Figure 1 is a block diagram of a multistage pulse amplifier with a low duty cycle; Figure 2 is a schematic illustration of an amplifier stage with an NMOS input; and Figure 3 is a schematic illustration of an alternative embodiment of the amplifier stage. The pulse amplifier with low duty cycle shown in Figure 1 consists of several cascaded fully differential, i.e. differential in/differential out, amplifier stages 1, a biasing network 2, and an offset canceling network 3 consisting of a low pass filter 4 and an ordinary differential amplifier 5. The gain of the amplifier 5 and the cut-off frequency of the low pass filter 4 in the offset canceling network should be chosen so that the resulting high pass characteristic of the canceling network has a cut-off frequency below the frequencies contained in the input signal applied to the amplifier. Figure 2 illustrates an amplifier stage 1 with a NMOS input, although it will be understood that the amplifier can be configured in opposite logic using PMOS components. Differential input signals IN+, IN-, are applied to respective NMOS transistors M1, M2 having a common node CN and together forming a differential pair. Transistors M1, M2 are in series with respective cascode transistors M5, M6 providing respective differential outputs OUT+, OUT-. Current sources I 1 , I 2 for the differential pair are connected to supply rail V DD . Load transistors M3 and M4 are connected in parallel with respective series combinations M1, M5 and M2, M6 to the same supply V DD as the current sources I 1 , I 2 for the differential pair M1, M2 in order to minimize the supply voltage needed to drive the amplifier. The two matching current sources I 1 and I 2 are larger than current source I, connected to common node CN to guarantee that there is always a current flowing through M3 and M4 even when the differential stage is fully saturated. This reduces the output voltage swing during clipping. It is possible to place the load transistors M3 and M4 in series with the respective series combinations M1, M5 and M2, M6, in which case the current sources should be placed where the respective load transistors are located in the parallel configuration. The two cascode transistors M5 and M6, which are provided between the drains of the differential pair and the two outputs OUT+, OUT-, minimize duty cycle errors for large input signals. The gate voltage to the cascode transistors is chosen to set the Vds of the differential pair above, but not substantially above, the saturation voltage of these transistors. This holds true as long as the amplifier circuit is not clipping due to large input signals. The cascode transistors ultimately turn off if a large input signal charges the common source node of the differential pair toward the output dc level so this node will be held relatively constant for very large input signals resulting in significantly more symmetric delays in positive and negative directions. In the presence of large input signals, one of the transistors of the differential pair, say M1, is driven hard into conduction, and its drain in effect gradually becomes shorted to the common node. The drain of transistor M1 is connected to the source of transistor M5, and consequently the voltage of the source of cascode transistor M5 approaches the voltage of its gate, so that the current flow through transistor M5 gradually reduces until it eventually turns off. This of course has the effect of gradually reducing the current flow through the associated differential transistor M1. In a typical design the common node will change 75mV to 150mV before the limiting circuit is activated. The gain is set by the ratio of transconductance in M1 and M3. It is advantageous to use the same type of transistors in both the differential pair and the load to minimize the gain variations due to process spread and the threshold voltage variations. In all normal cases the gain is larger than one, so M3 will have a lower transconductance than M1. With I 1 and I 2 being than I 3 , the dc current in the load transistors M3 and M4 will be larger than in M1 and M2, so W/L for M3 and M4 has to be smaller than in M1 and M2 to move them deeper into strong inversion and lower the transconductance. This results in a larger Vgs for the load transistors than for the differential pair, so there will be enough voltage over the current source for the differential pair to function correctly if the input of one gain stage is dc coupled to the output of the previous stage. The bias network generates the bias voltages for the current sources and the cascode transistors. With an ac coupled signal, the bias network also supplies the dc bias for the input node. In cases where the amplifier is powered on and off, the start-up time can be reduced if the offset cancellation feedback signal is held at a voltage near the dc input voltage while the amplifier is disabled. The current sources I 1 to I 3 can be implemented as simple current mirrors or cascoded current mirrors if better PSRR (Power Supply Rejection Ratio) is required. In an alternative embodiment, if, for instance, the gain is low or the output is ac coupled, the offset voltage at the output may be acceptable without offset cancellation. In this case the reference input is connected to a fixed voltage instead, typically the same voltage biasing the signal input. The circuit can be turned inverted with all NMOS transistors replaced by PMOS transistors and vice versa. The amplifiers can also be made variable gain amplifiers if the current source I 3 is made variable. A lower I 3 current will reduce the transconductance in the differential pair and also increase the current in the load transistors and increase their transconductance. If the dc level at the outputs of the amplifier stage must be set at a different voltage than slightly above the NMOS threshold, the NMOS load transistors M3, M4 can be replaced by PMOS transistors with their sources tied to their outputs, the drains to Vss and their gates to a voltage V G , which is approximately the desired output voltage minus a PMOS threshold voltage. This arrangement is shown in Figure 3. The drawback with this solution is a larger parameter variation due to process spread. The pulse amplifier is general in nature and can be used in a wide range of applications. Signals should be ac coupled since the input common mode range is limited. In particular, it can amplify a Manchester coded data stream at 300 kbits/sec. The amplitude ranges from 5 mV pp up to the supply voltage and maintains a duty cycle error of less than 5%. Advantages of the amplifier according to the present invention thus include: low duty cycle errors at high input levels due to well controlled clipping; low voltage operation by connecting the gain transistor and load transistor to the same supply rail; and the gain is relatively insensitive to temperature and process variations by the use of the same type of transistor as gain element and load. As mentioned above, the load transistors M3 and M4 can be placed in series with the cascode transistors M5, M6, in which case they change places with the current sources I 1 , I 2 . Although embodiments of the invention have been described above, it is not limited thereto and it will be apparent to those skilled in the art that numerous modifications form part of the present invention insofar as they do not depart from the scope of the claimed invention.

Claims: A pulse amplifier comprising an input (IN) for receiving input signals, an output (OUT), and a plurality of cascaded fully differential amplifier stages (1) between said input and output, wherein each said amplifier stage (1) comprises first and second transistors (M1, M2) forming a differential pair and having a common node (CN), and a pair of cascode transistors (M5, M6) in series with said respective first and second transistors (M 1,M2), each of said transistors (M1, M2, M5, M6) having first and second main electrodes and a gate electrode, the gate electrodes of said first and second transistors receiving input signals (IN+, IN-) for the amplifier stage (1), the second main electrodes of said first and second transistors (M1, M2) being connected to the first main electrodes of said respective cascode transistors (M5, M6), characterized in that said amplifier (1) further comprises a first series combination comprising a first load transistor (M3) in series with a first current source (I1) and having a first common connection point, and a second series combination comprising a second load transistor in series with a second current source (12) and having a second common connection point, said first and second common connection points providing output signals for said amplifier stage and being connected to the second main electrodes of said respective cascode transistors, and wherein in the presence of large input signals one of the transistors (M1 or M2) of said differential pair is driven hard into conduction causing the first main electrode of its associated cascode transistor (M5 or M6) to be substantially shorted to said common node (CN), thereby causing the voltage of the first main electrode of the associated cascode transistor (M5 or M6) to approach its gate voltage, and in turn causing a gradual reduction in the current flow through said associated cascode transistor (M5 or M6) and consequently a gradual reduction in the current flow through said one (M1 or M2) of the transistors of the differential pair (M1, M2) that was driven into hard conduction. A pulse amplifier as claimed in claim 1, wherein said first and second current sources (I1, I2) are connected in series with said series combinations of said first and second transistors (M1, M2) and their associated cascode transistors (M5, M6), and said first and second load transistors (M3, M4) are connected in parallel with said series combinations of said first and second transistors (M1, M2) and their associated cascode transistors (M5, M6). A pulse amplifier as claimed in claim 1, wherein said first and second current sources (I1, I2) are connected in parallel with said series combinations of said first and second transistors (M1, M2) and their associated cascode transistors (M5, M6), and said first and second load transistors (M3, M4) are connected in series with said series combinations of said first and second transistors (M1, M2) and their associated cascode transistors (M5, M6). A pulse amplifier as claimed in claim 1, wherein the gates of the cascode transistors (M5, M6) are connected to a voltage source (V B ) selected to set the source-drain voltage of said differential pair slightly above their saturation voltage, as long as the amplifier stage is not clipping due to large input signals. A pulse amplifier as claimed in any of claims 1 to 4, wherein said first main electrode is the source, and said second main electrode is the drain. A pulse amplifier as claimed in claim 5, further comprising a third current source (13) in series with said common node. A pulse amplifier as claimed in claim 6, wherein the currents generated by said first and second current sources (I1, I2) are greater than the current generated by said third current source (I3) to ensure that current flows through said load transistors when said first and second transistors are saturated. A pulse amplifier as claimed in claim 7, further comprising a biasing network for generating bias voltages for said current sources (I1, I2, I3) and said cascode transistors (M5, M6). A pulse amplifier as claimed in claim 8, wherein the load transistors (M3, M4) are of the same type as said differential transistors. A pulse amplifier as claimed in any preceding claim, further comprising an offset canceling network (4, 5) connected to said input. A pulse amplifier as claimed in claim 10, wherein said offset canceling network comprises an amplifier (5) and a low pass filter (4) having a cut-off frequency below the frequencies contained in said input signal. A pulse amplifier as claimed in claim 7, wherein said amplifier stages (1) have variable gain, and said third current source (I3) is variable. A pulse amplifier as claimed in any preceding claim, wherein said current sources (I1, I2, I3) are current mirrors. A pulse amplifier as claimed in any of claims 1 to 13, wherein said current sources (I1, I2, I3) are cascoded current mirrors.

Inventors: ANDERSSON BENGT-OLOV OLLE, ANDERSSON, BENGT-OLOV (OLLE)
Applicants: ZARLINK SEMICONDUCTOR AB
Application year: 1999
Main classification: H03K5
IPC codes: H03K5