SINGLE CHIP CMOS TRANSMITTER/RECEIVER AND VCO-MIXER STRUCTURE

Abstract: A single chip RF communication system and method and a VCO-mixer (130) structure are provided. The RF communication system in accordance with the present invention includes a transmitter (1100) and a receiver (100), an antenna for receiving transmitting RF signals, a PLL (130) for generating multi-phase clock signals having a frequency different from a carrier frequency in response to the multi-phase clock signals and a reference signal having the carrier frequency, a demodulation-mixing unit (140) for mixing the received signal with the multi-phase clock signals having a frequency different from the carrier frequency to output the RF signals having a frequency reduced by the carrier frequency and an A/D converting unit (160) for converting the RF signals from the mixing unit into digital signals. The VCO (130) in accordance with the present invention includes a plurality of differential delay and the mixer includes a differential amplifying circuit (1200A) and combining circuit (1200B). The differential amplifying circuit (1200A) of the multi-phase mixer includes two load resistors (R2, R1) coupled to two differential amplifiers (1200A1, 1200A2) respectively. The combining circuit (1200B) includes bias transistors (1232, 1234), first and second combining circuits coupled to the bias transistors, respectively and a current source coupled to the first and second combining units.

Description: SINGLE CHIP CMOS TRANSMITTER/RECEIVER AND VCO-MIXER STRUCTURE BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a communication system, and in particular, to a CMOS radio frequency (RF) communication system. The present invention also relates to a voltage controlled oscillator (VCO) and mixer, and more particularly, to a multi-phase VCO and mixer. 2. Background of the Related Art Presently, a radio frequency (RF) communications system has a variety of applications including PCS communication and IMT systems. As such, a CMOS chip integration of the system has been pursued to reduce the cost, size and power consumption. Generally, the RF communication system is composed of a RF front-end block and a base-band digital signal processing (DSP) block. Currently, the base-band DSP block can be implemented with low cost and low power CMOS technology. However, the RF front-end block cannot be implemented by CMOS technology because of limitations in speed and noise characteristics, which are below the speed and noise specification of popular RF communication systems. For example, the PCS hand-phone systems operate at a frequency over 2.0 GHz, but current CMOS technology reliably operates only up to approximately 1.0 GHz in terms of speed and noise. Hence, the RF front-end block is implemented using bipolar or bi-CMOS technology that has better speed and noise characteristics than CMOS technology, but is more expensive and consumes more power. Currently, two different types of RF architecture called "direct conversion" and "double conversion" are used for CMOS RF communication systems. Both architectures have advantages and disadvantages in terms of CMOS implementations. Fig. 1 shows a related art direct conversion CMOS RF communication system 100, including an antenna 105, a RF filter 110, a low noise amplifier (LNA) 120, a first mixer 140, a second mixer 145, a phase-locked loop (PLL) 130, a first low pass filter (LPF) 150, a second LPF 155, a first analog/digital (A/D) converter 160, a second A/D converter 165, a third mixer 160 and a power amplifier 170. The antenna 105 receives RF signals, and the selected RF signals are then filtered at the RF filter 110. The filtered RF signals are amplified with a gain at the LNA 120, and the RF signals passing through the LNA 120 are directly demodulated into base band signals by quadrature multiplication at the first and second mixers 140, 145. The PLL 130 preferably generates two types of clock signals, I signals and Q signals, using a voltage controlled oscillator (VCO). The I clock signals and the Q clock signals are the same, excepting a phase difference. I signals preferably have a phase difference of 90 degrees from Q signals. That is, Q signals are phase shifted with respect to quadrature phase shift I signals. The two sets of signals I, Q are preferably used to increase the ability of the RF system to identify or maintain received information regardless of noise and interference. Sending two types of signals having different phases reduces the probability of information loss or change. A demodulation frequency f 0 in Figure 1 is equal to a modulation frequency f 0 . The demodulated based band signals have a frequency reduced by the frequency f 0 from an original frequency to pass through the first and second LPF 150, 155 and eventually become respective signals required for A/D conversion at the first and second A/D converters 160, 165. The digital signals are then transferred to a baseband discrete-time signal processing (DSP) block (not shown). Channel selection is performed by changing frequency f 0 in at the phase-locked loop (PLL) 130. One of the possible causes for the approximately 1 GHz limitation on the reliability of CMOS technology is the structure of the VCO and the mixer in the PLL 130. Figure 2 shows a circuit diagram of a background VCO-mixer, wherein the VCO 10 includes four differential delay cells 12, 14, 16 and 18 and has a structure similar to a ring oscillator. The four delay cells 12, 14, 16, 18 are serially connected and generate a clock signal LO+ and an inverted clock signal LO-, each having a frequency f 0 . A control circuit for the VCO 10 that generates a frequency control signal includes a phase frequency detector 4, a charge pump 6 and a loop filter 8 that outputs the frequency control signal to each of the delay cells 12, 14, 16, 18. The phase frequency detector 4 receives a reference clock signal f ref and a VCO clock signal f vco from a reference clock divider circuit 2 and a VCO clock divider circuit 3, respectively. The frequency f 0 of the clock signals LO + and LO- is represented by M/K (f ref ) = f 0 Thus, the frequency f 0 is based on the reference clock signal f ref and the divider circuits 2, 3. The mixer 20, for example, a Gilbert - Multiplier, multiplies the input signals, such as radio frequency (RF) signals RF + and RF-, with the clock signals LO + and LO-. The mixer 20 includes two load resistors Rl, R2 coupled to a source voltage V DD , eight NMOS transistors 21-28, and a current source I S1 . The gates of the NMOS transistors 21, 22 are coupled to receive the clock signal LO + , and the gates of the NMOS transistors 23, 24 are coupled to receive the inverted clock signal LO-. The gates of the NMOS transistors 25, 26 receive a common bias voltage V Bιas . The gates of the NMOS transistors 27, 28 receive the RF signals RF + , RF-, respectively. Therefore, the clock signals LO + , LO- are multiplied with the RF signals RF + , RF- only when the transistors 25, 27 or the transistors 26, 28 are transitted to the "ON" state together. The output signals OUT + , OUT- of the mixer 20 have a frequency lower than its original frequency by the frequency f 0 of the clock signals LO + , LO-. " While a wide frequency range and a low phase noise are desirable for various applications, the VCO-mixer structure 10, 20 can only support up to a frequency of approximately 1 GHz with reliable phase noise and frequency range. The performance of the VCO-mixer structure 10, 20 deteriorates in terms of phase noise and frequency range, and is unacceptable as the frequency of the clock signals LO + , LO- from the VCO increases. Hence, the VCO 10 and the mixer 20 cannot be readily implemented when the frequency f 0 of the clock signals LO + , LO- exceeds approximately 1 GHz. As described above, the related art direct conversion RF system 100 has advantages for CMOS RF integration because of its simplicity. In the related art direct conversion RF system, only a single PLL is required and high-quality filters are not required. However, the related art direct conversion architecture has disadvantages that make single chip integration difficult or impossible. As shown in Figure 3A, clock signals cos Gύ LO t from a local oscillator (LO) such as the VCO may leak to either the mixer input or to the antenna, where radiations may occur because the local oscillator (LO) is at the same frequency as the RF carriers. The unintentionally transmitted clock signals Δ(t)cos ( LO t signals can reflect off nearby objects and be "re-received" by the mixer. The low pass filter outputs a signal M(t) + Δ(t) because of leakages of clock signals. As shown in Figure 3B, self-mixing with the local oscillator results in problems such as time variations or "wandering" DC-offsets at the output of the mixer. Figure 3B illustrates time variations and a DC-offset. "A" denotes a signal before the mixer and "B" denotes a signal after the mixer. The time-varying DC-offset, together with inherent circuit offsets, significantly reduce the dynamic range of the receiver portion. In addition, a direct conversion RF system requires a high-frequency, low-phase-noise PLL for channel selection, which is difficult to achieve with an integrated CMOS voltage controlled oscillator (VCO), for at least the reasons discussed above. Figure 4 shows a block diagram of a related art RF communication system 300 according to a double conversion architecture that considers all of the potential channels and frequency transistors. The RF communication system 300 includes an antenna 305, a RF filter 310, a LNA 320, a first mixer 340, a second mixer 345, and a first LPF 350, a second LPF 355, second stage mixers 370-373, a first adder 374, and a second adder 375. The RF communication system 300 further includes a third LPF 380, a fourth LPF 385, a first A/D converter 390, a second A/D converter 395, first and second PLLs 330, 335, a third mixer 360 and a power amplifier 370. The mixers 340, 345, 370-373 are all for demodulation, while the third mixer 360 is for modulation. The first and second mixers 340, 345 are for a selected RF frequency and the second stage mixers 370-373 are selected for an intermediate frequency (IF). The first PLL 330 generates clock signals at a high frequency or the RF frequency, the second PLL 335 generates clock signals having a low frequency or the intermediate frequency (IF). Transmission data are multiplied with the clock signals having the RF frequency from the PLL 330 to have a frequency reduced by the RF frequency from an original transmission data frequency. The output signals of the third mixer 360 are amplified with a gain at the power amplifier 370 and then radiated through the antenna 305 for transmission. For reception data, the antenna 305 receives RF signals and the RF filter 310 filters the RF signals. The filtered RF signals are amplified by the LNA 320 and converted into IF signals by the quadrature mixers 340, 345 with a single frequency local oscillator, generally a VCO. The PLL 330 generates clock signals for I signals and Q signals of the RF signals. The first mixer 340 multiplies the RF signals with the clock signals for the I signals having the RF frequency, and the second mixer 345 multiplies the RF signals with the Q signals having the RF frequency. The LPFs 350, 355 are used at an IF stage (i.e., first stage) to remove any frequency components not converted upon conversion to the IF signals, which allows all channels to pass to the second stage mixers 370-373. All of the channels at the IF stage are then frequency- translated directly to base-band frequency signals by the tunable PLL 335 for channel selection. Demodulated base band signals C pass low pass filters (LPF) 380, 385 and are converted into digital data by A/D converters 390, 395. The digital data is then transferred into a base-band discrete-time signal processing (DSP) block (not shown). As described above, the related art double conversion RF system 300 has various advantages. The related art double conversion RF system 300 performs the channel tuning using the lower-frequency, i.e., IF, second PLL 335, but not the high-frequency, i.e., RF, first PLL 330. Consequently, the high-frequency RF PLL 330 can be a fixed- frequency PLL that can be more effectively optimized. Further, since channel tuning is performed with the IF PLL 335, which operates at a lower frequency, the contribution of phase noise into channel selection can be reduced. However, the related art double conversion RF system 300 has various disadvantages. The related art double conversion RF system 300 uses two PLLs, which are difficult to integrate in a single chip. Further, the frequency of first PLL remains too high to be implemented with CMOS technology, and in particular, with a CMOS VCO. The structure of the VCO and mixer imposes an approximately 1 GHz limitation on the reliability of the CMOS technology. In addition, a self-mixing problem still occurs because the second PLL is at the same frequency of the IF desired carrier. Figure 5A illustrates leakage of clock signals in the RF communication system 300, and Figure 5B illustrates time variation and "wandering" DC-offset due to leaking clock signals Δ(t)cosQ L02 (t) (e.g., self-mixing) in the RF communication system 300 of Figure 4. In Figure 5A, the first mixer multiplies the RF signals with clock signals cosO L01 t for RF having a frequency C0 LO1 and outputs the RF signals with M(t)cos ( LO2 t having a frequency reduced by the frequency 0 LO1 . The second mixer multiples the RF signals from the first mixer with clock signals cosG LO2 for IF having a frequency O L02 . However, since the frequency of the output signals of the second mixer is same as the frequency of desired RF carriers before the LPFs. Thus, the output signals of the second mixer may leak to a substrate or may leak to the second mixer again. The time-varying DC-offset, together with inherent circuit offsets, significantly reduces the dynamic range of the receiver portion. The above references are incorporated by reference herein where appropriate for appropriate teachings of additional or alternative details, features and/or technical background. SUMMARY OF THE INVENTION An object of the present invention is to at least substantially obviate problems and disadvantages of the related art. A further object of the present invention is to fabricate a CMOS RF front end and method for using same that allows one chip integration of an RF communication system. Another object of the present invention is to provide an RF communication system and method with reduced cost and power requirements. Still another object of the present invention is to provide a reliable high speed, low noise CMOS RF communication system and method for using same. Another ob j ect of the present invention is to increase a frequency range of a RF front end of an RF communication system. A further object of the present invention is to fabricate a VCO-mixer on a single substrate. Another object of the present invention is to increase the frequency range of a VCO-mixer structure. Still another object of the present invention is to reduce the noise of a VCO- mixture structure. Another object of the present invention is to increase a performance of the VCO-mixer structure. To achieve at least the above objects and advantages in a whole or in parts and in accordance with the purpose of the present invention, as embodied and broadly described, the structure of the invention includes a receiving unit that receives signals, including selected signals having a carrier frequency, a PLL that generates multi-phase clock signals having a frequency different from a carrier frequency and a reference signal having the carrier frequency, and a demodulation-mixing unit that mixes the received signals with the multi-phase clock signals to output the selected signals having a frequency reduced by the carrier frequency. To further achieve the objects in a whole or in parts, in accordance with the purpose of the present invention, a single chip RF communication system includes a transceiver for receiving and transmitting RF signals, a PLL for generating 2N-phase clock signals having a frequency 2*f/N smaller than a carrier frequency, wherein N is a positive integer as a phase number and f 0 is the carrier frequency, a demodulation mixing unit for mixing the RF signals from the transceiver with 2N-phase clock signals from the PLL to output the RF signals having a frequency reduced by the carrier frequency and comprising a plurality of two input mixers, and a A/D converting unit for converting the RF signals from the demodulation mixing unit into digital signals. To still further achieve the objects in a whole or in parts, in accordance with the purpose of the present invention, a method of operating a RF communication system includes receiving signals including selected signals having a carrier frequency, generating multi-phase clock signals having a frequency different from the carrier frequency, and a reference signal having the carrier frequency, and mixing the received selected signals with the multi-phase clock signals to output the selected signals having a frequency reduced by the carrier frequency. To achieve the advantages and in accordance with the purpose of the present invention, as embodied and broadly described, the structure of the invention comprises a clock generator that generates a plurality of first clock signals having different phases, each first clock signal having first frequency less than a reference frequency, and a mixer coupled to said clock generator for receiving the plurality of first clock signals to generate a plurality of second clock signals having a second frequency which is substantially same as the reference frequency, wherein said mixer multiplies the plurality of second clock signals with input signals to provide output signals. Additional advantages, objects, and features of the invention will be set forth in part in the description which follows and in part will become apparent to those having ordinary skill in the art upon examination of the following or may be learned from practice of the invention. The objects and advantages of the invention may be realized and attained as particularly pointed out in the appended claims. BRIEF DESCRIPTION OF THE DRAWINGS The invention will be described in detail with reference to the following drawings in which like reference numerals refer to like elements wherein: Figure 1 is a circuit diagram showing a related art RF communication system; Figure 2 is a circuit diagram of a related art VCO-mixer structure; Figure 3 A is a diagram showing clock signal leakage in the circuit of Figure 1; Figure 3B is a diagram showing "self mixing" in the circuit of Figure 3A; Figure 4 is a circuit diagram showing another related art RF communication system; Figure 5A is a diagram showing clock signal leakage in the circuit of Figure 4; Figure 5B is a diagram showing "self mixing" in the circuit of Figure 5A; Figure 6 is a diagram showing a first preferred embodiment of a multi-phase, low frequency (MPLF) RF communication system according to the present invention; Figure 7 is a block diagram showing an exemplary PLL circuit; Figure 8 is a block diagram showing a receive portion of a RF communication system according to another preferred embodiment of the present invention; Figure 9 is a block diagram showing the RF communication system of Figure 8 with six phases; Figure 10 is a block diagram showing a receive portion of a RF communication system according to yet another preferred embodiment of the present invention; Figure 11 is a block diagram showing the RF communication system of Figure 10 with six phases; Figure 12 is a block diagram showing a transmit portion of a RF communication system according to still yet another preferred embodiment of the present invention; Figure 13A is a block diagram showing an exemplary VCO-mixer structure; Figure 13B is a circuit diagram showing the VCO-mixer structure of Figure 13A; Figure 14 is a circuit diagram showing another exemplary VCO-mixer; and Figures 15A-15H are diagrams showing operational timing waveforms of Figure 14. DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS A single chip radio frequency (RF) communication system formed using CMOS techniques has various requirements. A CMOS voltage controlled oscillator (VCO) has poor noise characteristics. Accordingly, a CMOS phase-locked loop (PLL) integration is required. However, the number of PLL should be small and the center frequency of a PLL preferably differs sufficiently from a transmitting RF frequency (e.g., preferably low enough) to control a phase noise result using the CMOS VCO. High-quality filters are preferably eliminated because of associated disadvantageous area and power specifications. Also, a number of components in the CMOS RF system should be small or reduced without performance degradation. A first preferred embodiment of the present invention is a "multi-phase, low frequency" (MPLF) conversion RF communication system 500 shown in Figure 6 and can preferably be formed on a single CMOS chip. The first preferred embodiment can operate at frequencies well above approximately 1 GHz. The phrase "multi-phase low frequency conversion" is used because a single-phase periodic signal having a high frequency is preferably obtained by multiplying multi-phase low-frequency periodic signals. The first preferred embodiment of the MPLF conversion RF communication system 500 includes a front-end MPLF RF block 502 and a digital signal processing (DSP) block 504, which is preferably base-band. As discussed above, related art DSP blocks can be formed of CMOS techniques. Accordingly, a detailed explanation of the DSP block 502 including a digital signal processor 550 will be omitted. The MPLF conversion RF block 502 includes an antenna 505, an RF filter 510 (e.g., band pass filter), a low noise amplifier (LNA) 520 and first and second mixers 530, 560, respectively. The MPLF conversion RF block 502 further includes a phase- locked loop (PLL) 540, a low pass filter (LPF) 580, an analog/digital (A/D) converter 590 and a power amplifier 570 coupled between the second mixer 560 and the antenna 505. The PLL 540 generates a modulating and de-modulating clock, i.e., local oscillator (LO), whose frequency is determined by a reference clock (REF f 0 ). Figure 7 shows a block diagram of an exemplary embodiment of the PLL 540. The PLL 540 includes reference and main dividers 610, 620, respectively, a phase comparator 630, a loop filter 640 and a voltage controlled oscillator (VCO) 650. The VCO 650 outputs the LO frequency f 0 , which is compared to the reference clock signal by the phase comparator 630. An output signal of the phase comparator 630 is passed though the loop filter 640 as a control signal (e.g., frequency) for the VCO 650. The frequency of the LO is preferably varied according to the communication system. For example, the LO frequency for a personal communication system (PCS) can be about 1.8 GHz, and the LO frequency for the IMT 2000 system is about 2.0 GHz. In the first preferred embodiment of the MPLF conversion RF communication system 500 shown in Figure 6, transmission data is received by the MPLF RF block 502 from the DSP block 504. The transmission data is modulated by a preferably modulating second mixer 560 at the LO frequency. The modulated data is amplified by the power amplifier 570 and output by the antenna 505. The low noise amplifier (LNA) 520 receives an input signal from the antenna 505 and amplifies the signal level to output an RF signal. The RF BPF 520 is preferably coupled between the antenna 505 and the LNA 520. The RF signal is demodulated by the de-modulating first mixer 530 at preferably the same frequency as the modulation frequency. The output of the de-modulating mixer 530 becomes received data by passing the LPF 580. The received data is preferably converted to a digital signal by the A/D converter 590 and output to the DSP 550. In order to use a single PLL with a center frequency sufficiently lower than a transmitting RF frequency, the first preferred embodiment of the MPLF conversion RF communication system 500 uses a single-phase high-frequency periodic signal (i.e., RF frequency) obtained by multiplying a multi-phase low-frequency periodic signal together. In particular, a high frequency "sine" and "cosine" signal is needed in a RF system, although the present invention is not intended to be so limited. Sine and cosine signals, which have frequencies of ( >^, can be obtained by multiplying N-phase sine signals that have frequencies of KJ^/N as shown in equations 1 and 2: (1) (2) A multiplication factor is not "N" but "N/2" because the remaining N/2 sine signals can be an inverted version of the first N/2 sine signals. The inverted signals are preferably used to make differential signals for a differential input mixer. Figure 8 shows a receive portion 700 of a second preferred embodiment of a RF block according to the present invention, which can be used in the first preferred embodiment of the MPLF conversion RF communication system. The receive portion 700 includes an antenna 715, an RF filter 720, a LNA 725 and a demodulation mixer 730. The receive portion 700 of the RF block further includes a PLL 740, a low pass filter 780 and an analog/digital converter 790. The PLL 740 generates a de-modulating clock, i.e., local oscillator (LO) equal to 2*f 0 /N, whose frequency is determined by a reference clock (not shown). The antenna 715, the RF filter 720, the LNA 725, the LPF 780 and the analog/digital converter 790 operate similar to the first preferred embodiment, and accordingly, a detailed explanation is omitted. The receive portion 700 of the RF block uses one PLL 740. The PLL 740 uses a frequency of 2*f 0 /N, and generates in total 2N-phase clock signals. The PLL 740 generates N-phase χLO C0S (k,t) and N-phase ±LO sm (k,t) signals, which are preferably determined as shown in equations 3-4. (3) 2ω RF 2kπ π N Λ ±LO (k,ή = ± sin(- 1- + — )where,k- 0,1,2... — - 1 N N N 2 ±LD . (k,t) = ± άn( —t- ^ -)ιι>here,k= 0,1,2,...— - I (4) ^ N N 2 As shown in Figure 8, the receive portion 700 of the RF block has the demodulating mixer 730 divided into upper and lower mixer arrays 732, 734. Each of the upper and lower mixer arrays 732, 734 includes a plurality of conventional 2-input mixers 735. The upper mixer array 732 multiplies N-phase (N/2: un-inverted, N/2: inverted) with a frequency of (2Q RF )/N, sine signals and a RF signal, which is equivalent to multiplying single phase, frequency of O^, cosine signals and the RF signal. Both un-inverted and inverted sine signals are needed for inputting to a single mixer because the conventional 2-input mixer requires differential input. The lower mixer array 734 multiplies N-phase (N/2 non-inverted, N/2 inverted) with a frequency of Gj R p/N, sine signals and the RF signal, which is equivalent to multiplying single phase, frequency of Cύ^ sine signals and the RF signal. Thus, the receive portion 700 of the RF block functions equivalently with the direct conversion architecture shown in Figure 1. However, the receive portion 700 according to the present invention uses the N-phase, a frequency of 2ω RF /N, sine signals in de-modulation in contrast to the single phase, and a frequency of 0)^ sine signals. As described above, the PLL 740 generates 2N-phase clock signals. N-phase clock signals are N-phase sine signals and N-phase cosine signals. Both the N-phase signals includes N/2 non-inverted signals and N/2 inverted signals. The N-phase sine signals are input to the upper mixer array 732 together with the RF signals, and the N- phase sine signals are input to the lower mixer array 734, together with the RF signals. The upper and lower mixer arrays 732 and 734 have a plurality of mixers 735 and a M number of stages respectively. The M number of stages includes a first stage, (e.g., 735), a second stage (e.g., 735 ' ),..., a M-lth stage, and a Mth stage (e.g., 735 "). Each stage of each mixer array includes at least one mixer having two inputs. The number Kl of mixer at the first stage is the highest number of stages. The last stage, the Mth stage, has the lowest number (KM) of mixers among the whole stages. The relative order of the mixer-number among the stages may be expressed the inequality K1>K2>K3)K4 KM-1>KM. Each mixer 735 has two inputs. Each input has an inverted signal and a non- inverted signal of the inverted signal because each input of the mixers 735 inputs two different signals. As described above, the RF signals from the LNA 725 and the N- signals from the PLL 746 are used as the input signals of mixers 735 at a first stage. Output signals of mixers 735 at the first stage are used as input signals of mixers 735 at the second stage. In a same manner, output signals of mixers at the M-lth stage are used as two input signals of a mixer 735 ", which is a single mixer at the Mth stage of the upper mixer array 732 and the lower mixer array 734. Figure 9 shows a 6-phase example for the receive portion 700 of an MPLF conversion RF communication system that uses the conventional 2-input mixer. A PLL 840 generates 12-phase sine signals, which are transmitted to a mixer 830. The phase difference between adjacent two signals is 7l/6 (i.e., 271/12). Phases (0,2,4,6,8,10) are used as inputs to an upper mixer 832 and multiplied together with the preferably RF input, which is equivalent with multiplying cos (Oϋ R pt) and the RF input. Phases (1,3,5,7,9,11) are input to a lower mixer 834 and multiplied together with the preferably RF input, which is equivalent with multiplying sin (CO^t) and the RF input. Accordingly, the frequency of the clock signals is f 0 when the clock signals are multiplied with the RF signals. The PLL 840 includes a clock generator such as a voltage controlled source (VCO) and thus generates 12-phase clock signals for the multiplication with the RF signals upon demodulation. The generated clock signals have a frequency 2*f 0 /P (P=phase number) lower than a frequency f 0 to be multiplied with the RF signals. The clock signals from the PLL 840 may have the lower frequency 2*f 0 /P because the PLL 840 generates multi-phase clock signals phase 0, , phase 12. Filtered RF signals are amplified with a gain in the LNA 725 and multiplied with the multi-phase clock signals, resulting in 12 sine signals in the mixer array 830 for modulation. The RF signals multiplied with the clock signals have a frequency lower than an original frequency by a final frequency f 0 of the clock signals. The initial frequency 2 f 0 /P of the clock signals from the PLL 840 is changed to the frequency f 0 for multiplication with the RF signals in the mixer (e.g., mixer array) 830. Therefore, the upper mixer array 832 and the lower mixer array 834 combine the clock signals having the frequency 2*f 0 /P and multiply the clock signals having frequency f 0 with the RF signals. Consequently, the RF signals having a frequency reduced by frequency f 0 pass through the LPFs 780 and the A/D converters 790 and are sent to a DSP part (not shown). The 12 phase sine signals generated by the PLL 840 are shown as follows: ω Phase 0: sin ( H t+ Ξ, ω Phase 1: sin ( Rf t) ω Phase 2 : sin (■ 3 6 Phase 3: sιn( 1- ) Phase 4: sin 1- — ) 3 6 P mh , ase 5 r.: sm . ( , ω R 1,- — 4π. ) 3 6 P .h,ase r 6: -sin i RF 1.+ , — π 3 6 Cύ RF Phase 7: -sin( ) 3 Phase 8 : -sin( t- — ) 3 6 Phase 9 : ■ sin ( - ) 3 6 ω Phase 10 RF 3π sin ( ■ ω Rf π Phase 11 : - sin ( 1- ) 3 6 Figure 10 shows a MPLF conversion receive portion 900 of an RF block according to a third preferred embodiment of the present invention, which can be used in the first preferred embodiment of the MPLF conversion RF communication system. The receive portion 900 includes an antenna 915, a RF filter 920, a LNA 925 and mixer 930. The receive portion 900 of the RF block further includes a PLL 940, a LPF 90 and an A/D converter 990. The PLL 940 preferably generates a de-modulating clock, i.e., local oscillator (LO) preferably equal to 2*f RF /N, whose frequency is determined by a reference clock (not shown). The antenna 915, the RF filter 920, the LNA 925, the LPF 980 and the A/D converter 990 operate similar to the first preferred embodiment, and accordingly, a detailed explanation is omitted. The receive portion 900 of the RF block uses just one PLL. The PLL 940 includes a clock generator 942 preferably using a frequency of 2 >;" f RF /N. The clock generator 942 preferably generates N-phase χLO C0S (k,t) and N-phase ±LO sin (k,t) signals, which total 2N phase signals. The clock generator 942 is preferably a multi- phase VCO and the mixing section 930 is also a multi-phase mixer. As shown in Figure 10, the receive portion 900 of the RF block uses multi-phase mixers 932 and 934. The upper multi-phase mixer 932 replaces the function of the upper mixer array 732 and the lower multi-phase mixer 934 replaces the function of the lower mixer array 734. The PLL 940 can generate clock signals for modulation and demodulation. The clock generator 942 of the PLL 940 generates clock signals having a frequency 2*f 0 /N (N = phase number) for demodulation and modulation. The clock generator 942 generates clock signals with frequency 2*f 0 /N because of frequency limits according to CMOS device implementation. For a CMOS implementation of a RF communication system, a frequency of the clock generator 942 should be different and lower than that of the mixing section 930. Figure 11 shows a 6-phase example of a receive portion 1000 of an MPLF conversion RF communication system that uses a multi-phase input mixer. A PLL 1040 generates 12-phase sine signals, which are transmitted to a multi-phase mixer 1030. Phases (0,2,4,6,8,10) are used as inputs to an upper mixer 1032 and multiplied together with a preferably RF input, which is equivalent with multiplying cos (CO j ^t) and the RF input. Phases (1,3,5,7,9,11) are input to a lower mixer 1034 and multiplied together with a preferably RF input, which is equivalent with multiplying sin (CO^t) and the RF input. Figure 12 shows a MPLF conversion transmit portion 1100 of an RF block according to a fourth preferred embodiment of the present invention, which can be used in the first preferred embodiment of the MPLF conversion RF communication system. The receive portion 1100 includes an antenna 1105, a mixer 1160, a PLL 1140, a plurality of LPFs 1180, a plurality of D/A converters 1190 and a power amplifier 1170 coupled between the mixer 1160 and the antenna 1105. The PLL 1140 generates clock signals using a clock generator 1142. The clock generator 1142 preferably generates a modulating and de-modulating clock signal using a local oscillator (LO), whose frequency is determined by a reference clock (f^). In the fourth preferred embodiment of the transmit portion 1100 of an RF block, digital data is received from a DSP block (not shown) and converted into an analog signal by the D/A converter 1190 and filtered by the LPF 1180. The mixer 1160 preferably receives multi-phase low frequency (i.e., 2*f 0 /N) clock signals from the PLL 1140 and a base band signal from the LPF 1180 to generate a modulated RF signal whose frequency is The mixer 1160 preferably includes multi-phase up conversion mixers 1165. Figure 12 also shows a block diagram of an exemplary embodiment of the multi-phase up conversion mixer 1165. The mixer 1165 uses two control circuit blocks 1162 and 1164, which receive the clock signals LO(0, ..., N-l), /LO(0, ..., N-l), to generate the modulated RF signal. The modulated RF data is amplified by the power amplifier 1170 and is then output by the antenna 1105. As described above, a mixer for demodulation reduces a high frequency of RF signals received with a frequency of clock signal by multiplying the RF signals with the clock signals. In the fourth preferred embodiment, the mixer 1160 preferably modulates the transmission data to increase a low frequency of the transmission data by a frequency of the combined clock signals. Noise does not effect the transmission data as significantly for modulation as it does for demodulation. However, reducing the frequency of the clock signals LO(0, ..., N-l) does reduce or remove noise such as parasitic capacitance. In addition, the frequency limit of the CMOS technology of approximately 1 GHz can be overcome. Thus, the fourth preferred embodiment has the same advantages as the first through third preferred embodiments. Figure 13A is a block diagram of an exemplary VCO-mixer structure in accordance with a preferred embodiment of the present invention. The VCO-mixer circuit is described in U.S. Patent Application No. 09/121,863, entitled "VOC-MIXER STRUCTURE" by Mr. Kyeongho Lee, the subject matter of which is hereby incorporated by reference. The structure includes a multi-phase voltage controlled oscillator VCO 1250 and a multi-phase mixer 1200. The multi-phase mixer 1200 includes a differential amplifying circuit 1200A and a combining circuit 1200B. When a reference clock signal having used, the multi-phase VCO 1250 generates a plurality of N-phase clock signals LO(i = 0 to N-l) having a frequency of 2*f 0 /N, where N = N D *2 and N D equals the number of delay cells in the multi-phase VCO 1250. In other words, the VCO 1250 reduces the frequency f 0 to 2*f 0 /N, thus reducing the phase noise of the multi-phase VCO and increasing the frequency range. The plurality of N-phase intermediate clock signals LO(0), LO(l),....,LO(N-l) having a frequency of 2*f 0 /N is inputted into the combining circuit 1200B of the multi- phase mixer 1200, and the input signals, for example, RF signals RF + , RF- are inputted into the differential amplifying circuit 1200A. The differential amplifying circuit 1200B differentially amplifies the radio frequency signals RF + , RF-. The combining circuit 1200B is responsive to a bias voltage V Bιas and combines the N-phase intermediate clock signals LO(0)-LO(N-1) to generate the output clock signals LOT + , LOT- having the original frequency f 0 . The mixer 1200 then accomplishes a multiplication of the output clock signals LOT + , LOT- and the RF signals RF + , RF-. Figure 13B illustrates an exemplary circuit diagram of the VCO-mixer structure 1250, 1200. The multi-phase VCO 1250 includes N D number of delay cells 1250 1 -1250 ND coupled in series. Based on that configuration, the multi-phase VCO generates a plurality of N-phase intermediate clock signals LO(0)-LO(N-1) having a frequency of 2 f 0 /N. A control circuit for the VCO 1250 that generates a frequency control signal includes a phase frequency detector 1254, a charge pump 1256 and a loop filter 1258 that outputs the frequency control signal to each of the delay cells ^O^^O NE ,. The phase frequency detector 1254 receives a reference clock signal f ref and a VCO clock signal f vco from a reference clock divider circuit 1252 and a VCO clock divider circuit 1253, respectively. The frequency 2*f< N of the clock signals LO(φ)-LO(N-l) is represented by M7K'(f ref ) = 2f 0 /N. Thus, the frequency f 0 is based on the reference clock signal f ref and the divider circuits 1252, 1253. In other words, f vco can be 2f 0 /N setting M'/K' of the divider circuits 1252, 1253. The differential amplifying circuit 1200A of the multi-phase mixer 1200 includes two load resistors Rl', R2' coupled to two differential amplifiers 1200A 1; 1200A 2 , respectively. The first differential amplifier 1200A, includes two NMOS transistors 1210, 1212, and the second differential amplifier 1200A 2 also includes two NMOS transistors 1214, 1216. The drains of the NMOS transistors 1210, 1216 are coupled to the load resistors Rl', R2', respectively, and the gates of the NMOS transistors 1210, 1216 are coupled for receiving the RF signal RF + . Further, the drains of the NMOS transistors 1212, 1214 are coupled to the load resistors R2', Rl', respectively, and the gates are coupled for receiving the RF signal RF-. The sources of NMOS transistors 1210, 1212 and NMOS transistors 1214, 1216 are coupled to each other, and to the combining circuit 1200B of the multi-phase mixer. The differential amplifiers 1200A 1; 1200A 2 differentially amplify the RF signals RF + , RF-, respectively, such that more accurate output signals OUT- , OUT+ can be obtained. Further, the differential amplification removes noise that may have been added to the RF signals RF + , RF-. In the present preferred embodiment, two differential amplifiers 1200A J 1200A 2 are included. However, the present invention may be also accomplished using only one of the differential amplifiers in alternative embodiments. The combining circuit 1200B includes bias NMOS transistors 1232, 1234, first combining unit 120QiS> 1 and second combining unit 1200B 2 coupled to the bias NMOS transistors 1232, 1234, respectively, and a current source I sl . coupled to the first and second combining units 12006^ 1200B 2 . The first combining unit 1200B J includes a plurality of transistor units 1220 0 , 1220 2 ...1220 N . 2 , and the second combining unit includes a second plurality of transistor units 1220,, 1220 J ...1220 N . ! . Preferably, each of the plurality of transistor units includes a plurality of serially connected transistors, wherein the serially connected transistors are coupled in parallel with the serially connected transistors of the plurality of transistor units. Preferably, each transistor unit includes two (2) serially connected transistors. Hence, in the preferred embodiment, there are a total of N/2 number of transistor units in each combining unit 1200A or 1200B, such that the total number of NMOS transistors is 2*N. The gate of the bias NMOS transistors 1232, 1234 are coupled for receiving the bias voltage V Bιas , and the gates of the transistors in the first and second plurality of transistor units are coupled for receiving a corresponding N-phase intermediate clock signals LO(i) and /LO(i) having a frequency of 2*f 0 /N, where /LO(i) = LO(N/2 + i), i =0, 1.., N/2-1. In the present preferred embodiment, the bias NMOS transistors 1232, 1234 are included for prevention of error, however, such transistors may be omitted in alternative embodiments. Further, the sequential ON-OFF operation of the 2*N number NMOS transistors of the combining circuit 1200B is equivalent to a NAND logic circuit, which can be interchanged with other equivalent logic circuits and structure in alternative embodiments. The generic Figure 13B structure allows integration of the multi-phase VCO 1250 and multi-phase mixer 1200 on a single chip, i.e., on a single semiconductor substrate using CMOS technology. Such structure and layout reduce noise including noise caused by parasitic capacitances. As described above, the differential amplification using the RF signals RF + and RF- in the differential amplifying circuit 1200A reduces noise. The reduction of the reference frequency f 0 to N-phase intermediate clock signals LO(i) having a frequency of 2 f 0 /N also reduces noise. When a plurality of transistors are formed on the same substrate, such as a semiconductor substrate for CMOS technology, a plurality of P-N junctions are formed in the substrate. The parasitic capacitances mostly exist at the P-N junctions. If the frequency of a signal applied to the gate of the transistor is very high, the higher frequency of f 0 causes much more noise compared to a reduced frequency of 2 f 0 /N. Further, the operation of the differential amplifier circuit 1200A and the combining circuit 1200B is dependent on the output clock signals LOT + , LOT- having a frequency of f 0 , which are provided by the first and second combining units HOOB^ 1200B 2 , respectively, by combining the N-phase intermediate clock signals LO(i) having a frequency of 2*f 0 /N. When the bias voltage V Bιas is applied, the NMOS transistors 1232, 1234 are transited to the ON and OFF states based on the output clock signals LOT + , LOT-. Although the NMOS transistors 1210, 1212, 1214 and 1216 are transited to the ON state by the RF signals RF + , RF- applied to the gate electrodes, the amplification of the RF signals RF + , RF- and the output clock signals LOT+ , LOT- for generating the output signals OUT + , OUT- is performed when the bias NMOS transistors 1232, 1234 are turned on by the clock signals LOT+ , LOT-. Figure 14 illustrates another preferred embodiment of the multi-phase VCO and the multi-phase mixer when N D = 3 and N = 6, and Figures 15A-15H illustrate the operational timing diagrams of the circuit of the preferred embodiment illustrated in Figure 14. The multi-phase VCO 1250 includes three delay cells 1250 1 -1250 3 to generate 6-phase intermediate clock signals LO(0)-LO(5). An exemplary circuit including five transistors for the delay cells 1250 1 -1250 3 (i.e., the delay cell 1250^ is also shown. For illustrative purposes only, if the input clock signal has a frequency of f 0 = 1.5 GHz, the 6-phase intermediate clock signals LO(0)-LO(5) will have a frequency of 0.5 GHz. The 6-phase mixer 1280 includes a differential amplifying circuit 1280A and a combining circuit 1280B. The differential amplifying circuit 1280A includes a first differential amplifier 1280A, having NMOS transistors 1260 and 1262 and a second differential amplifier 1280A 2 having NMOS transistors 1264 and 1266, which are coupled to load resistors R3 and R4, respectively. The combining circuit 1280B includes a first and second combining unit 1280B ! 1280B 2 , which are commonly coupled to a current source I S2 . The first and second combining units 12806, ^ 1280B 2 are coupled to the first and second differential amplifiers ^δOA^ 1280A 2 through bias NMOS transistors 1282,1284, respectively, which are biased by a bias voltage V Bias . Cumulatively, the first and second combining units 12506^ 1250B 2 includes six transistor units 1270 0 -1270 5 with a total of twelve transistors. As shown in Figures 15A-15F, the 6-phase VCO 1250 generates 6-phase intermediate clock signals LO(l)-LO(5) having the reduced frequency f 0 /3. The 6-phase mixer 1250 receives the 6-phase intermediate clock signals LO(l)-LO(5) and the RF signals RF + and RF-. Each intermediate clock signal LO(l)-LO(5) and /LO(0)-/LO(2), where /LO(0) = LO(3), /LO(l) = LO(4) and /LO(2) = LO(5), is applied to a corresponding transistor of the first and second combining units 1280B 1; 1280B 2 The first and second combining units 1280B ! 1280B 2 combine the 6-phase intermediate clock signals LO(0), LO(l),...LO(4), LO(5) having the frequency f 0 /3 to generate the output clock signals LOT+ and LOT- having the frequency f 0 . When LO(0) is high and LO(l) is low (LO(4) = high), the two output signals LOT + , LOT- are low and high, respectively. When LO(l) is high and LO(2) is low (LO(5) = high), the output signals LOT + , LOT- are high and low, respectively. When LO(2) is high and LO(3) is low (LO(0) = high), the output signals LOT + , LOT- are low and high, respectively. When LO(3) is high and LO(4) is low (LO(l) = high), the output signals LOT + , LOT- are high and low, respectively. When LO(4) is high and LO(5) is low (LO(2) = high), the output signals LOT + , LOT- of the mixer 503 are low and high, respectively. When LO(5) is high and LO(0) is low (LO(3) = high), the output signals LOT + , LOT- are low and high, respectively. Each pair of NMOS transistors in the combining circuit are turned on in order, thereby producing the output signals LOT + , LOT-, as shown in Figures 15G and 15H. As described above, the preferred embodiments have various advantages. The preferred embodiment of the MPLF conversion RF communication system does not need any high quality filter and uses just one PLL. Thus, the MPLF conversion architecture can be easily integrated in one CMOS chip. Further, the frequency of channel selecting PLL is reduced from F R to βf j ^/N, which results in the reduction of phase noise of a clock generating circuit such as a VCO and easy implementation of channel selection. In particular, the PLL frequency (LO) is different from (e.g. smaller than) the carrier frequency. As a result, the preferred embodiments of the MTLF RF communication system includes at least the advantages of both the related art direct conversion and double conversion communication systems while eliminating disadvantages of both architectures. Additionally, a robust and low noise CO and mixer can be fabricated on a single substrate, preferably on a semiconductor substrate, using CMOS technology. The interference caused by the input signal and the input clock signal is drastically reduced, because the frequency of the intermediate clock signals deviate from the modulation frequency. The phase locked loop (PLL) frequency range can be increased, because the PLL frequency range can be easily increased on the low-center frequency condition. Moreover, such results can enhance the channel selection capability of RF front-end in a RF communication system. The foregoing embodiments are merely exemplary and are not to be construed as limiting the present invention. The present teaching can be readily applied to other types of apparatuses. The description of the present invention is intended to be illustrative, and not to limit the scope of the claims. Many alternatives, modifications, and variations will be apparent to those skilled in the art. In the claims, means-plus- function clauses are intended to cover the structures described herein as performing the recited function and not only structural equivalents but also equivalent structures.

Claims: WHAT IS CLAIMED IS: 1. A communication system, comprising: a receiver unit that receives signals including selected signals having a carrier frequency; a phase-locked loop that generates multi-phase clock signals having a frequency different from the carrier frequency and a reference signal having the carrier frequency; and a demodulation-mixing unit that mixes the selected signals received by the receiver unit with the multi-phase clock signals to output the selected signals having a frequency reduced by the carrier frequency. 2. The communication system of claim 1, wherein the frequency is smaller than the carrier frequency, and wherein the carrier frequency is greater than about 1GHz, and further wherein the phase-locked loop includes a clock generator. 3. The communication system of claim 1, wherein the receiver unit is a transceiver, further comprising : a modulation mixer that mixes the multi-phase clock signals with transmission data to modulate the transmission data; and a power amplifier that amplifies the modulated transmission data and transmits the data to the transceiver for transmission. 4. The communication system of claim 1, further comprising: a RF filter coupled to the receiver unit that filters the selected signals received by the receiver unit; a low noise amplifier coupled to the RF filter that amplifies the selected signals filtered by the RF filter with a gain; a low pass filter coupled to the demodulation-mixing unit that filters the selected signals having the frequency reduced by the carrier frequency; an A/D converting unit that converts the selected signals from the mixing unit into digital signals; and a discrete-time signal processing unit that receives the digital signals. 5. The communication system of claim 1, wherein: the communication system is a RF receiver portion; the selected signals are RF signals; the multi-phase clock signals have a frequency of (2 >:" carrier frequency /N), wherein N is a positive integer; and the RF communication system is formed on a single CMOS chip. 6. A single chip RF communication system, comprising: a transceiver for receiving and transmitting RF signals; a phase-locked loop for generating 2N-phase clock signals having a frequency 2*f 0 /N smaller than a carrier frequency, wherein N is a positive integer as a phase number and f 0 is the carrier frequency; a demodulation mixing unit for mixing the RF signals from the transceiver with 2N-phase clock signals from the phase-locked loop to output the RF signals having a frequency reduced by the carrier frequency, wherein the demodulation mixer comprises a plurality of two input mixers; and an A/D converting unit for converting the RF signals from the demodulation mixing unit into digital signals. 7. The communication system of claim 6, wherein the demodulation mixing unit comprises a first mixer array comprising one half of the two input mixers and a second mixer array comprising the other half of the two input mixers, wherein the first and second mixer array inputs each corresponding N-phase clock signal of the 2N-phase clock signals together with the RF signals. 8. The communication system of claim 6, wherein each mixer array comprises multi-stages of mixers, each stage comprising at least one two-input mixer, and a first stage of the multi-stages inputs the RF signals and N-phase clock signals. 9. The communication system of claim 8, wherein the multi-stages have a corresponding reducing number of mixers Kl > K2 > K3 > > Ki, where Kl is the first stage, K2 is a second stage, K3 is a third stage, and Ki is a ith stage. 10. A method of operating a RF communication system, comprising: receiving signals including selected signals having a carrier frequency; generating multi-phase clock signals having a frequency different from the carrier frequency, and a reference signal having the carrier frequency; and mixing the received selected signals with the multi-phase clock signals to output the selected signals having a frequency reduced by the carrier frequency. 11. The method of claim 10, further comprising : RF filtering the received selected signals; amplifying the filtered selected signals with a gain; low pass filtering the selected signals having the frequency reduced by the carrier frequency; A/D converting the low pass filtered frequency reduced selected signals into digital signals; and discrete-time signal processing the digital signals. 12. The method of claim 10, further comprising: modulation mixing the multi-phase clock signals with transmission data to modulate the transmission data; and power amplifying the modulated transmission data and transmitting the data to the transceiver for transmission. 13. A circuit comprising: a clock generator that generates a plurality of first clock signals having different phases, each first clock signal having a first frequency which is less than a reference frequency; and a mixer coupled to said clock generator for receiving the plurality of first clock signals to generate a plurality of second clock signals having a second frequency which is substantially same as the reference frequency, wherein said mixer multiplies the plurality of second clock signals with input signals to provide output signals. 14. The circuit of claim 13, wherein said clock generator includes a plurality of delay cells coupled in series for providing the plurality of first clock signals having different phases. 15. The circuit of claim 13, wherein said mixer includes: a differential amplifying circuit for receiving the input signals and providing the output signals; and a combining circuit for receiving the plurality of first clock signals from said clock generator. 16. The circuit of claim 15, wherein said differential amplifying circuit includes: at least one load resistor coupled for receiving a first potential; and at least one differential amplifier coupled to one of the said load resistors, and the combining circuit. 17. The circuit of claim 15, wherein said combining circuit comprises: a first combining unit for receiving corresponding first clock signals and coupled to said differential amplifying circuit to output a corresponding second clock signal; a second combining unit for receiving corresponding first clock signals and coupled to said differential amplifying circuit to output a corresponding second clock signal; and a current source coupled to said first and second combining units and coupled for receiving a second potential. 18. The circuit of claim 17, further comprising first and second bias transistors respectively coupled between said first and second combining units and the differential amplifying circuit, wherein each of said first and second combining units comprises a plurality of transistor units coupled to each other in one of series and parallel.

Inventors: LEE KYEONGHO, LEE, KYEONGHO
Applicants: GLOBAL COMMUNICATION TECHNOLOG, GLOBAL COMMUNICATION TECHNOLOGY, INC.
Application year: 1999
Main classification: H03K9
IPC codes: H03D1, H03D7, H03H11, H03K9, H03L7, H04B1, H04L7, H04L27