LINE DRIVER WITH ADAPTIVE OUTPUT IMPEDANCE

Abstract: A line driver comprising a first transistor (M1), a first operational transconductance amplifier (A1) and a reference resistor (10) for converting an input voltage (Vin) to a first current (i1) through the first transistor (M1). A second current i2=n*i1 flows through a second transistor (M2) which forms a 1:n current mirror with the first transistor (M1). The current i2 flows to a load (6), if so required via a transmission line (TL). The impendance of the load (6) is equal to the characteristic impedance RL of the transmission line (TL). Thus, the impedance seen by the line driver is equal to RL. A second operational transconductance amplifier (A2) counteracts reflected signals in the output signal (Vout) caused by mismatch between the output impedance of the current mirror (M1, M2) and the impendance seen by the line driver.

Description: Line driver with adaptive output impedance The invention relates to a line driver, more particularly to a line driver with adaptive output impedance. Such a line driver is known from the International Patent Application published under No. WO 95/02931. A line driver is an electronic buffer amplifier designed to have an output impedance matched to the characteristic impedance of a transmission line. Transmission lines are widely used for conveying electric signals. To minimise reflections the source and load impedances of the transmission line should be equal to the characteristic impedance of the transmission line. A standard value for video applications is 75 ohms. A buffer amplifier designed to drive a 75 ohm transmission line should have an output impedance of 75 ohms in order to minimise reflections. The buffer sees a load resistance of 75 ohms, i.e. the impedance of the terminated transmission line. Figure 1 shows a first known approach to implement such a buffer. The buffer provides a low-impedance voltage source VS with a series resistor Rs having a value equal to the characteristic impedance of the transmission line TL in order to implement the correct output impedance of the line driver. This type of line driver is often used in digital-to-analog converters (DAC) where the output voltage is controlled by a digital input signal. An disadvantage of this first known structure is that the voltage source VS has to deliver twice the desired output voltage. This becomes a problem where the available supply voltage drops while the signal levels remain unchanged; e.g. a 1.5Vpp output voltage at 3V supply voltage is barely possible. Figure 2 shows a second known approach. The voltage source with series resistance is replaced by a current source CS with a parallel resistance Rs. In this structure no voltage is lost, but half of the current delivered by the current source CS is wasted in the parallel resistance Rs. This technique is often used in DACs where the output current is digitally controlled. It appears that both known solutions are not attractive since either voltage or current is wasted. Therefore it can be appreciated that a line driver with specified output impedance which matches the impedance of a transmission line and which does not require twice the output voltage or twice the output current is desirable. It is therefore an object of this invention to provide a line driver with specified output impedance without the voltage loss inherent in the series resistor approach or without the current loss inherent in the parallel resistor approach. According to the invention there is provided a line driver comprising: a line driver input terminal for receiving an input signal, - a line driver output terminal for connecting a load, a current mirror having a current gain n, comprising: a first transistor having a main current path inserted between a voltage supply terminal and a reference node, and a second transistor having a main current path inserted between the voltage supply terminal and the line driver output terminal, respective control electrodes of the first transistor and second transistor being connected to receive substantially the same control voltage, a reference resistor coupled to the reference node and having a resistance substantially equal to n times the characteristic resistance of the load, a first operational transconductance amplifier having differential input terminals coupled to the line driver input terminal and to the reference node, and having an output terminal coupled to the respective control electrodes of the first transistor and the second transistor, and a second operational transconductance amplifier having differential input terminals coupled to the line driver input terminal and to the liner driver output terminal, and having an output terminal coupled to the output terminal of the first operational amplifier. Due to the gain of the first operational transconductance amplifier (OTA) the voltage at the reference resistor is substantially equal to the input voltage. The input signal is converted to a current through the reference resistor. An n times amplified current flows through the load. Since the resistance of the reference resistor is n times the resistance of the load the voltage across the load is equal to the input voltage. No signal voltage is wasted. The supply voltage can be low and should allow only for a proper voltage drop across the second transistor. Moreover, hardly any signal current is wasted, except for a relatively small current which flows through the first transistor and the reference resistor. By making the current gain n of the current mirror sufficiently high, for example n = 10 to n=40 the current waste is negligible. The second OTA senses any unwanted reflected voltages at the line driver output terminal and converts these reflected voltages to a current which flows from the output terminal of the second OTA into the output terminal of the first OTA. The current flowing into the output terminal of the first OTA in turn generates a differential voltage at the differential input terminals of the first OTA, which voltage has the same amplitude but the reverse sign as the reflected voltage. The reverse reflected voltage is added to the input voltage and thus counteracts the reflected voltage, and the line driver behaves like a source with an output impedance equal to the load impedance. The linearity of the line driver may be increased in an embodiment which further comprises a first bias current source coupled to the reference node, for supplying a first bias current, and a second bias current source coupled to the line driver output, terminal for supplying a second bias current having a value which is substantially equal to n times the value of the first bias current. The accuracy of the resistance of the reference resistor may be insufficient in certain integrating processes. If this lack of accuracy is a problem the line driver may further comprise a first bias current source coupled to the reference node, for supplying a first bias current, and a second bias current source coupled to the line driver output terminal, for supplying a second bias current having a value which is substantially equal to n times the value of the first bias current, or alternatively it may comprise means for adjusting the current gain n of the current mirror in response to a low-frequency voltage difference between the line driver input terminal and the line driver output terminal. Instead of the voltage difference between the line driver input terminal and the line driver output terminal the voltage difference between the reference node and the line driver output terminal may be used for adjusting the resistance of the reference resistor or for adjusting the current gain of the current mirror. These and other aspects of the invention will be described in more detail, by way of example, with reference to the accompanying drawings, in which: Figure 1 shows a circuit diagram of a conventional line driver with voltage source and series resistor; Figure 2 shows a circuit diagram of a conventional line driver with current source and parallel resistor; Figure 3 shows a circuit diagram of a first embodiment of a line driver according to the invention, Figure 4 shows a circuit diagram of a second embodiment of a line driver according to the invention, Figure 5 shows a circuit diagram of a third embodiment of a line driver according to the invention, and Figure 6 shows a part of a circuit diagram of an alternative to the third embodiment of Figure 5. In the Figures corresponding elements have the same reference signs. Figure 3 shows the circuit diagram of an embodiment of a line driver according to the invention. The line driver has a line driver input terminal 2, which receives an input signal Vin. The input signal Vin may be a video signal, an audio signal, a telephone signal, a digital data signal etc., provided with an appropriate DC bias. The line driver further has a line driver output terminal 4 for connecting a load 6 via a transmission line TL. The transmission line, however, is optional and can be omitted if so desired. The resistance RL of the load 6 is equal to the characteristic impedance of the transmission line TL, for example 75 ohms. As is known from the art of transmission lines, the impedance seen at the line driver output terminal 4 is equal to the characteristic impedance RL of the transmission line TL. The line driver further comprises a reference node 8 to which a reference resistor 10 is connected. The reference resistor 10 has a resistance Rl which is n times the resistance RL of the load 6, i.e. Rl =n*RL, n being a positive number. The main current path of a first PMOS transistor Ml is inserted between the reference node 8 and a positive supply terminal 12 and the main current path of a second PMOS transistor M2 is inserted between the line driver output terminal 4 and the positive supply terminal 12. The control electrodes or gates of the transistors Ml and M2 are interconnected and receive the same control voltage. The transistors Ml and M2 thus form a current mirror or current amplifier, i.e. there is a fixed ratio (current gain) between the current through the first transistor Ml and the current through the second transistor M2. By a suitable design, for example by proportioning the geometries of the first and second transistors Ml and M2 in the ratio l:n, the current ratio can be fixed at the same factor n as mentioned before in relation to the resistances of the reference resistor 10 and the load 6. A typical value for the current gain n is in the range from 10 to 40. A first operational transconductance amplifier (OTA) Al has its non-inverting input terminal 14 coupled to the reference node 8, and has its inverting input terminal 16 coupled to the line driver input terminal 2. The output terminal 18 of the first OTA Al drives the interconnected control electrodes of the transistors Ml and M2. The first OTA Al has a transconductance gm, which means that the current supplied at output terminal 18 is equal to gm times the voltage difference between the non-inverting input terminal 14 and the inverting input terminal 16. Assuming that the gain in the system of OTA Al and transistor Ml is adequate, the signal voltage at the reference node 8 is substantially equal to the input voltage Vin at the line driver input terminal 2. The input voltage Vin thus causes a current il =Vin/Rl to flow through the reference resistor 10. Since Rl =n*RL this current is also equal to il =Vin/(n*RL). Due to the current amplification n the current i2 through transistor M2 is n times the current through transistor Ml: i2=n*il =Vin/RL. This means that the output voltage Vout at the line driver output terminal 4 is also equal to Vin. No signal voltage is wasted between the input voltage Vin at the line driver input terminal 2 and the output voltage Vout at the line driver output terminal 4. The positive supply voltage at the positive supply terminal 12 should be high enough to allow a proper voltage drop across the transistors Ml and M2. A supply voltage as low as 3V is sufficient to drive 1.5Vpp in a 75 Ohm load. The output impedance of the second transistor M2 is high and would not match with the characteristic impedance RL of the terminated transmission line TL. A second OTA A2 is added to emulate the correct output impedance RL at the line driver output terminal 4. The second OTA A2 is preferably assumed to be equal to the first OTA Al and preferably has the same transconductance gm. The second OTA A2 has its inverting input terminal 20 coupled to the line driver input terminal 2 and its non-inverting input terminal 22 to the line driver output terminal 4. The output terminal 24 of the second OTA A2 is coupled to the output terminal 18 of the first OTA Al. If no reflections occur at the line driver output terminal 4, i.e. when Vout=Vin, the differential input voltage for the second OTA A2 is zero and the second OTA A2 has no further effect. In the case of reflections, however, the second OTA A2 comes into action. Assuming that the reflected voltage is equal to dVout, the output voltage is now Vout +d Vout and the second OTA A2 sees a differential voltage dVout at its input terminals 20 and 22. In response to this voltage difference the second OTA A2 generates an output current gm*dVout, which can only flow into the output terminal 18 of the first OTA Al. Owing to the feedback around the first OTA Al the current imposed at output terminal 18 causes a differential voltage -dVout at the input terminals 14 and 16 of the first OTA Al. The voltage across the reference resistor 10 is now Vin -d Vout and the resulting current il through the reference resistor 10 is copied with a gain n to the line driver output terminal 4. The output current 12 decreases with (n*dVout)/(n*RL)=dVout/RL. The line driver thus has an output impedance equal to the characteristic impedance RL. Figure 4 shows a modified version of the line driver of Figure 3. A first bias current source 26 is connected to the reference node 8 and provides a bias current Idc through the first transistor Ml. Similarly, a second bias current source 28 is connected to the line driver output terminal 4 and provides a bias current n*Idc through the second transistor M2. The addition of the two bias current sources 26 and 28 enhances the linearity of the line driver, while slightly more power is consumed. Figure 5 shows an embodiment in which the reference resistor is an electronically variable resistor RV with a resistance control terminal 30 for varying the resistance of the reference resistor 10. The resistance control terminal 30 is driven by an output 32 of a differential amplifier A3, which has one of its inputs, for example a non- inverting input 34, connected to receive the input voltage Vin and the other input, for example an inverting input 36, connected to receive the output voltage Vout. The amplifier A3 preferably has a small bandwidth and tunes the DC resistance Rl of the variable resistor RV to the desired value n*RL. The tuning may be useful where the accuracy of the reference resistor 10 is a problem due to process variations or where a load with a different resistance is connected. The non-inverting input 34 of amplifier A3 may alternatively be connected to receive the voltage at the reference node 8 instead of at the line driver input terminal 2, as indicated in Figure 5 with a dashed line. Instead of by tuning the resistance of the reference resistor 10, the current gain n can be tuned by making the current gain of the current mirror M1/M2 variable. Figure 6 illustrates a possible solution for making a variable current gain n. A variable resistor RV2 is connected in series with the source of transistor Ml. The control terminal 40 of the variable resistor RV2 is connected to the output 32 of the differential amplifier A3. The input terminals 34 and 36 of the differential amplifier A3 are connected as shown in Figure 5. Again the alternative connection to the reference node 8 is indicated with a dashed line. A further resistor 42 may be connected in series with the source of transistor M2 for reasons of symmetry. The invention is not limited to the embodiments shown in the Figures. Instead of unipolar MOS transistors bipolar transistors may be employed, in which case the base, emitter and collector of a bipolar transistor replace the gate, source and drain of a unipolar transistor. The current mirror configuration can be a more sophisticated or elaborated one. Resistors may be inserted in series with the sources of the transistors Ml and M2; cascode transistors may be inserted in series with the drains of the transistors Ml and M2 etc. In principle, any current mirror configuration with a current gain n can be used for this purpose, but it will be appreciated that complex current mirror configurations generally need more supply voltage to operate properly. More generally, a line driver is disclosed comprising a first transistor Ml, a first OTA Al and a reference resistor 10 for converting an input voltage Vin to a first current il through the first transistor Ml. A second current i2=n*il flows through a second transistor M2, which forms a l:n current mirror with the first transistor Ml. The current i2 flows to a load 6 via a transmission line TL. The transmission line TL, however, is optional. The impedance of the load 6 is equal to the characteristic impedance RL of the transmission line TL. Thus, the impedance seen by the line driver is equal to RL. A second OTA A2 counteracts reflected signals dV in the output signal Vout caused by mismatch between the output impedance of the current mirror Ml, M2 and the impedance seen by the line driver.

Claims: Claims 1. A line driver comprising: a line driver input terminal (2) for receiving an input signal, a line driver output terminal (4) for connecting a load (TL, 6), a current mirror (Ml, M2) having a current gain n, comprising: a first transistor (Ml) having a main current path inserted between a voltage supply terminal (12) and a reference node (8), and a second transistor (M2) having a main current path inserted between the voltage supply terminal (12) and the line driver output terminal (4), respective control electrodes of the first transistor (Ml) and second transistor (M2) being connected to receive substantially the same control voltage, - a reference resistor (10) coupled to the reference node (8) and having a resistance substantially equal to n times the characteristic resistance of the load (TL, 6), a first operational transconductance amplifier (Al) having differential input terminals (14, 16) coupled to the line driver input terminal (2) and to the reference node (8), and having an output terminal (18) coupled to the respective control electrodes of the first transistor (Ml) and the second transistor (M2), and a second operational transconductance amplifier (A2) having differential input terminals (20, 22) coupled to the line driver input terminal (2) and to the liner driver output terminal (4), and having an output terminal (24) coupled to the output terminal (18) of the first operational amplifier (Al). 2. A line driver as claimed in Claim 1, further comprising means (A3, 30, RV) for adjusting the resistance of the reference resistor (10) in response to a low frequency voltage difference between the line driver input terminal (2) and the line driver output terminal (4). 3. A line driver as claimed in Claim 2, wherein the means for adjusting comprises a variable resistor (RV) coupled to the reference node (8), and a differential amplifier (A3) having differential input terminals (34, 36) coupled to the line driver input terminal (2) and to the liner driver output terminal (4), and having an output (32) coupled to a resistance control terminal (30) of the variable resistor (RV). 4. A line driver as claimed in Claim 1, further comprising means (A3, 30, RV) for adjusting the resistance of the reference resistor (10) in response to a low frequency voltage difference between the reference node (8) and the line driver output terminal (4). 5. A line driver as claimed in Claim 4, wherein the means for adjusting comprises a variable resistor (RV) coupled to the reference node (8), and a differential amplifier (A3) having differential input terminals (34, 36) coupled to the reference node (8) and to the liner driver output terminal (4), and having an output (32) coupled to a resistance control terminal (30) of the variable resistor (RV). 6. A line driver as claimed in Claim 1, 2, 3, 4 or 5, further comprising means (A3, RV2, 40) for adjusting the current gain n of the current mirror in response to a low frequency voltage difference between the line driver input terminal (2) and the line driver output terminal (4). 7. A line driver as claimed in Claim 1, 2, 3, 4 or 5, further comprising means (A3, RV2, 40) for adjusting the current gain n of the current mirror in response to a low frequency voltage difference between the reference node (8) and the line driver output terminal (4). 8. A line driver as claimed in Claim 1, 2, 3, 4, 5, 6 or 7, further comprising a first bias current source (26) coupled to the reference node (8) for supplying a first bias current, and a second bias current source (28) coupled to the line driver output terminal (4) for supplying a second bias current having a value which is substantially equal to n times the value of the first bias current.

Inventors: NAUTA BRAM, NAUTA, BRAM
Applicants: KONINKL PHILIPS ELECTRONICS NV, PHILIPS NORDEN AB, KONINKLIJKE PHILIPS ELECTRONICS N.V.
Application year: 1998
Main classification: H04L25
IPC codes: H04L25