Method and device for estimating the frequency offset in a signal received by a mobile-telephone demodulator

Abstract: Device for estimating the frequency offset in a signal received by a differential demodulator of a mobile-telephone set. The device has means (1) for calculating an error signal ε on the basis of the phase  of the output signal of the demodulator and the ideal phase  o of this signal, means (4) for establishing the absolute value (ABS) of the error signal ε, means (2, 5, 7) for calculating an estimate of the average of the error signal ε calculated over a certain number of samples, from which estimate n FO-RAW bits are taken, means (3, 6, 8) for calculating an estimate of the average of the absolute value of the error signal ε over a certain number of samples, from which estimate n Q-RAW bits are taken, and means (10) for modelling a frequency-offset function Ω(FO-RAW, Q-RAW) on the basis of the said estimates.

Description: The present invention relates to a device for estimating the frequency offset, in particular for a mobile-telephone demodulator. Frequency modulation is the modulation technique most widely used in mobile radio systems. In frequency modulation, the signals have all their information contained in the phase or the frequency of the carrier. In order to recover the data, the demodulator which is used must estimate the phase or the frequency of the signal. If the carrier frequency is not exactly the expected frequency (because of a difference in frequency between the frequency of the oscillators of the transmitter and of the receiver), this may lead to an inaccuracy, and therefore increase the error rate of the demodulator. It will be assumed below that the oscillator of the receiver is perfect but the oscillator of the transmitter is imperfect. In reality, both are imperfect. One of the great problems with radio systems is that of reducing the impact of the frequency error of the received signal by compensating for it locally. Indeed, in order to reduce the total system cost, manufacturers are reluctant to use oscillators which are accurate and therefore expensive. There are usually two ways of compensating for the frequency error of the carrier: a) the frequency of the oscillator of the receiver is changed in order to adapt it to the frequency of the carrier. This involves external compensation using a VTCXO; b) the frequency is corrected directly inside the demodulator. This therefore involves internal compensation. In both cases, it is necessary to have a good estimate of the frequency error in a probably imperfect environment (low signal/noise ratio) in a fairly short time. Furthermore, if the receiver successively receives signals from different transmitters, it must frequently estimate new frequency errors. In this case, the frequency error estimator must give a reliable estimate on the basis of a very small number of samples. Document EP 0 533 208 A2 describes a method of coherently detecting a multi-phase PSK IF analog signal, comprising the steps of generating a digital baseband signal, multiplying said digital baseband by a local signal, applying the obtained signal to a plurality of single-tuned filters which have tuning frequencies staggered, these filters generating a resulting signal, a frequency error signal and a correlation coefficient, selecting one of the resulting signals, and generating a reproduced carrier by dividing this resulting signal. Document EP-0 252 500 describes an 8-phase phase-shift keying demodulator comprising a transversal equalizer including a transversal filter and a tap gain control signal generator, first means for coherently demodulating the equalized signal, second means for generating a plurality of error signals, third means for receiving the error signals and determining whether a signal point of said 8-phase phase modulated signal is usable for controlling the transversal equalizer, fourth means for detecting asynchronism of the demodulator, fifth means connected to the fourth means and controlling the tap gain control signal generator when said fourth means indicates asynchronism of the demodulator. Document EP-0 564 361 relates to a demodulation apparatus for a signal modulated on two quadrature axes by the pi/4-QPSK modulation technique alternately employing two phase-shifted constellations. This apparatus comprises a voltage-controlled oscillator providing a local signal having the carrier frequency, demodulation means providing the phase P and quadrature Q components of the demodulated received signal, phase locking means producing a control signal for controlling the oscillator. The phase locking means comprise means of alternate reversal of the polarity, in tempo with the alternating of the constellations, and providing the control signal for controlling the oscillator. Document EP-01 08358 describes a phase demodulator comprising a first phase detector receiving a modulated carrier wave, a variable phase shifter control by a control signal, a second phase detector receiving the modulated carrier waves and related to the variable phase shifter, A/D converters for multilevel discriminating outputs of the first and second phase detectors; and logic means for processing the output of the A/D converter and providing an orthogonality error signal as control signal. The paper "A comparison study of low bit rate DE-QPSK and TCM 8-PSK fully digital demodulators over a land mobile satellite link" presents the performance of different digital trellis coded (TMC) 8-PSK coherent demodulator operating over a mobile satellite channel. The paper "VLSI implemented 60 Mb/s QPSK/OQPSK burst digital demodulator for radio application" presents a fully digital high speed Quadrature Phase Shift Keying (QPSK) Offset QPSK (OQPSK) burst demodulator for radio applications, which has been implemented on a 0.5 µm Complementary Metal Oxide Semiconductor (CMOS) master slice Very Large Scale Integrated circuit (VLSI). The object of the invention is to create a very efficient frequency error estimator used with a differential phase demodulator. A further object of the invention is to create a frequency error estimator which is particularly well-suited to a π/4 QPSK demodulator, having four-state phase modulation with a jump of π/4, dedicated to wireless applications such as PHS and PDC mobile telephones. The invention therefore relates to a method for estimating the frequency offset in a signal received by a demodulator of a mobile-telephone set. The method being characterized by the feature of claim 1. The invention also relates to an estimation device according to claim 3. The invention will be understood more clearly with the aid of the following description, given solely by way of example and with reference to the appended drawings, in which: Fig. 1 is a diagram representing the phase points of a QPSK modulation; Fig. 2 is a diagram which is similar to that in Figure 1 and shows the impact of the frequency offset on the differential QPSK demodulation; Fig. 3 is a diagram representing the effect of the frequency error on the in-phase component I of the signal in π/4 QPSK modulation; Fig. 4 is a histogram of the signal I; Fig. 5 is a diagram representing the histogram of I in a noisy environment, without frequency error; Fig. 6 is a diagram representing the histogram of I in a noisy environment, with frequency error; Fig. 7 is a diagram representing the impact of the frequency offset on the quality estimate; and Fig. 8 is a block diagram of the circuit for estimating frequency offset according to the invention. In the following description, the impact of the frequency error on the value of the phase of the demodulated samples will firstly be explained. The principle of the invention will then be described. Lastly, the frequency error estimator used in PHS mobile telephone applications will be described. The influence of the frequency error will firstly be examined by studying the PSK modulation. In the case of phase modulation, the information is contained in the phase of the signal: U(t) = A cos(2πf o t + (t)) where U(t) is the modulated signal f o is the carrier frequency (t) is the phase of the signal, as a function of the information. At the sampling point, (kT) generally belongs to a certain group of possible values. Figure 1 shows four QPSK modulation phase points. The impact of the frequency error will now be considered. There are in principle two ways of demodulating the signal: a) multiplying the received signal with an internal frequency tuned to the expected carrier frequency (coherent demodulation), b) multiplying the received signal with the same signal stored one symbol in front (differential demodulation). In the case of coherent demodulation, the following is obtained: M(t)= A cos(2πf 1 t + (t))x B cos(2πf o t +  o ), where -M(t) is the first step in the demodulation -f o is the carrier frequency f 1 is the carrier frequency increased by a frequency error (f 1 = fo + f error ) (t) is the phase of the signal as a function of the information  o is the initial phase of the internal signal. It is assumed that  o = 0 in order to simplify the notation. M(t)= AB[cos(2π(f 1 +f 0 )t+(t))+cos(2π(f 1 -f 0 )t+(t))]= X+Z The first expression X is filtered in the demodulator. The second expression Z contains the information. When there is no error on the frequency of the transmitter, Z = cos((t)). When there is an error on the frequency of the transmitter Z = cos(2πf error t+(t)). In- the case of incoherent differential demodulation, the following equation is obtained: M(t)=A cos(2πf 1 t+(t))* B cos(2πf 1 (t+T) + (t+T)) in which M(t) is the first step in the demodulation, f 1 is the carrier frequency plus a frequency error (f 1 = f  +f error ), (t) is the phase of the signal, depending on the information, T is the period of the symbol: M(t)= AB[cos(2πf 1 (2t+T)+(t))+(t+T))+cos(2πf 1 T+(t+T)-(+))] = X+Z The first expression X is filtered in the demodulator. The second expression Z contains the information. When there is no error on the frequency of the transmitter, Z = cos(2πf 0 T+(t+T)-(t)). f 0 and T are customarily chosen so that T = nx1/f  . In this case, Z = cos((t)). When there is an error on the frequency of the transmitter: Z = cos(2πf error T+(t+T)-(t)) Z = cos( e +(t+T)-(t)) where  e is a constant phase error due to the frequency error. The impact of the frequency offset in the case of incoherent demodulation is thus a constant phase error. This phase error can be corrected easily. The role of the frequency offset error estimator is to give an estimate of  e as quickly as possible and with the best reliability, even in a highly unfavourable environment. The impact of the frequency offset on differential PSK demodulation is represented in Figure 2. The general use of a frequency offset estimator will now be described. A normal demodulator calculates the in-phase and quadrature components I and Q of the signal in order to recover the data. The effect of the frequency offset estimator on the signals I and Q will therefore be explained. However, the principle remains the same if the demodulator directly estimates the phase of the signal or another monotonic function of the phase. The example will be taken of π/4 QPSK modulation used in PHS, PDC, PACS systems, for example. The principle is still valid with other differential demodulations of the PSK type. The frequency offset estimator uses the error between the sample (white dot in Figure 2) and the ideal sample (grey dot). This error is referred to as ε. As can be seen clearly in Figure 3, in the case of estimating I and Q, it can be seen that the sign of the impact of a frequency offset depends on the quadrant in which the sample lies. If the sample is in the grey part, the error on I is +ε. If the sample is in the white part, the error is -ε. The effect of the frequency error on the component I is represented in Figure 3. It will be assumed below that the sign is compensated so that the impact of the frequency error is always ε irrespective of the sample. In the case of π/4 QPSK modulation, the histogram of I (or Q) with or without offset under perfect conditions is represented in Figure 4. This explains why a normal offset estimator uses the average of the error ε to obtain an estimate of the frequency error. This estimate will be referred to below as FO-RAW = average (ε). However, in the case of an unfavourable environment, for example in the presence of (additive) white gaussian noise (AWGN), the histogram of I becomes larger and demodulation errors occur, in particular in the case of a frequency offset error. Figure 5 shows the histogram of the in-phase component I in a noisy environment and without frequency error. When erroneous demodulation occurs, the error estimate ε also becomes erroneous. In Figure 6, the true error on the "dark" samples is ε1. However, because of the erroneous demodulation, the error estimate is ε2. Figure 6 shows the histogram of I in a noisy environment with a frequency error. Because of these errors, the estimate FO-RAW is less than that which is expected. The error on FO-RAW increases with the frequency error and the amount of noise. It can therefore be seen that the estimate FO-RAW depends on the frequency offset and also on the quality of the received signal. FO-RAW = FO-RAW (frequency offset, quality) The quality estimate will now be described. It can be seen that it is important to evaluate the quality of the signal in order to obtain a reliable frequency offset estimate. The usual way of estimating the quality of a signal is to evaluate the variance of the signal, that is to say to evaluate the average of the absolute value of the phase error, which is referred to as Q-RAW = average (ε). In this estimator, the impact of a demodulation error is not so serious because the absolute value is used. However, even in a highly favourable environment, the estimate Q-RAW is bound to increase with the frequency offset. In the case of a large frequency offset, Q-RAW delivers a high value even if the quality of the signal is very good. Figure 7 shows the impact of the frequency offset on the quality estimate. The result of what has gone before is that the estimator Q-RAW depends on the quality of the signal and also on the frequency error. Q-RAW = Q-RAW (frequency offset, quality). According to the invention, the frequency offset information contained both in the signals FO-RAW and Q-RAW is used, and the quality information is removed. To do this, it is necessary to evaluate the following function Ω. Ω (FO-RAW, Q-RAW) = frequency offset. An approximation of Ω is made by using a ROM memory. The ROM memory may, of course, be synthesized by combinatorial logic in order to obtain a more efficient embodiment. For programming the ROM memory, there are two ways of evaluating the function Ω, that is to say calculating the best coefficients to be stored in the ROM. Ω may be calculated directly using system equations. In this case, all that needs to be done is to quantify the function Ω according to the possible values of Q-RAW and FO-RAW. In most cases, it is difficult to define the function Ω explicitly, in particular when an attempt is made to take the effect of noise into account. The coefficients could be calculated by simulation or by experimentation. The best way is to fix the frequency error arbitrarily at a certain known value for each test n in a series of tests. Let δn be the frequency error during test number n. For this test number n, let FO-RAW n and Q-RAW n be the corresponding values of FO-RAW and Q-RAW. Ω (FO-RAW, Q-RAW) is defined by the average value of all the δ n such that FO-RAW = FO-RAW n and Q-RAW = Q-RAW n . In the case of a π/4 QPSK differential demodulator (four-state phase modulation with a jump of π/4), the invention is implemented as follows. The demodulator provides the frequency offset estimator with an estimate of the in-phase and quadrature components I and Q of the signal which it has received. These components I and Q are compared with the components I o and Q o of the ideal position. The error ε on I is then estimated and corrected in order to remove the variation in sign according to the value Q of the sample, as indicated at the start of the description. The estimates FO-RAW and Q-RAW are calculated over the amount of samples available. With only 12 samples, the embodiment obtains a very good result even in an unfavourable environment. n FO-RAW and n Q-RAW are respectively the number of bits used in FO-RAW and Q-RAW. n freq.offset is the number of bits in the estimate of the frequency offset. The total dimension of the ROM is: ROM dimension = 2 (n FO-RAW + n Q-RAW )*n freq.offset The ROM memory may be replaced by a logic circuit. In this case, the dimension of the embodiment may be improved considerably by virtue of the redundancy and the symmetry in Ω. The circuit for estimating frequency offset according to the invention is represented in Figure 8. It comprises an error calculation circuit 1 which, on its inputs, receives the signals I, Q drawn from the transmitted signal and I o , Q o corresponding to the ideal phase. The circuit 1 is connected at its output to a first adder 2 directly and to a second adder 3 via a circuit 4 for establishing the absolute value ABS of the error signal ε appearing at the output of the error calculation circuit 1. The output of the first adder 2 is connected to an FO-RAW circuit 5 intended to take n FO-RAW bits from the available FO-RAW bits. The output of the second adder 3 is connected to the input of a Q-RAW circuit 6 intended to take n Q-RAW bits from the available Q-RAW bits. The outputs of the first and second adders 2 and 3 are furthermore looped back onto the second inputs via respective one-symbol delay circuits 7, 8. The outputs of the circuits 5 and 6 are connected to corresponding address inputs of a ROM memory 10 which is programmed in the manner indicated above and which constitutes means for modelling the function Ω(FO-RAW, Q-RAW). The frequency offset signal appears at the output of this ROM memory. It should be noted that the same embodiment may be used to obtain a quality estimate independent of the frequency error. Certain results given by the circuit for estimating the frequency offset according to the invention are given below. In the case of the embodiment in Figure 8, the result of the frequency offset is given with an accuracy of 3 ppm. The application range is -9 ppm to 9 ppm. The method and the device for estimating frequency offset which has just been described is particularly well-suited to wireless applications characterized by a short learning sequence and an unfavourable communication environment. The algorithm firstly estimates the average of the phase error of the demodulated signal and its variance. By virtue of these two statistical parameters and the involvement of the ROM memory or combinatorial logic, a highly reliable frequency offset estimate is obtained, with means which have a very low degree of complexity. The efficiency of the method and of the device of the invention makes it possible to correct the frequency within a packet, and not to wait for the next data packet in order to apply the estimate. This characteristic is highly beneficial in the case of demodulating a plurality of base stations which have different frequency offsets.

Claims: Method for estimating the frequency offset in a signal received by a demodulator of a mobile-telephone set, characterized in that it comprises the steps of : calculating an error signal ε on the basis of the phase Φ of the output signal of the demodulator and of the ideal phase Φ o of this signal ; calculating estimates of the averages of the signal ε (FO-RAW) and of its absolute value (Q-RAW) over a certain number of samples, taking a certain number of bits (n FO-RAW , n Q-RAW ) of these estimates and applying them to means for modelling a function Ω (FO-RAW, Q-RAW) representing the frequency offset, wherein the modelling of the function Ω (FO-RAW, Q-RAW) comprises arbitrarily fixing the frequency error δn at a certain known value for each n of a series of tests, FO-RAW n and Q-RAW n being the corresponding values of FO-RAW and Q-RAW, and defining the function Ω (FO-RAW, Q-RAW) by the average value of all the errors such δn that FO-RAW = FO-RAW n and Q-RAW = Q-RAW n . Estimation method according to claim 1, wherein the output of the demodulator is formed by its in-phase component I and its quadrature component Q, and the error signal ε is calculated on the basis of these in-phase and quadrature components I and Q of the demodulated signal and of the in-phase and quadrature components I o and Q o corresponding to a signal having ideal phase. Device for estimating the frequency offset in a signal received by a differential demodulator of a mobile-telephone set, intended for implementing the method according to one of Claims 1 to 2, characterized in that it comprises means (1) for calculating an error signal ε on the basis of the phase  of the output signal of the demodulator and of the ideal phase  o of this signal, means (4) for establishing the absolute value (ABS) of the error signal ε, means (2, 5, 7) for calculating an estimate of the average of the error signal ε calculated over a certain number of samples, from which estimate n FO-RAW bits are taken, means (3, 6, 8) for calculating an estimate of the average of the absolute value of the error signal ε over a certain number of samples, from which estimate n Q-RAW bits are taken, and means (10) for modelling a frequency-offset function Ω(FO-RAW, Q-RAW) on the basis of the said estimates, wherein the modelling of the function Ω (FO-RAW, Q-RAW) comprises arbitrarily fixing the frequency error δn at a certain known value for each n of a series of tests, FO-RAW n and Q-RAW n being the corresponding values of FO-RAW and Q-RAW, and defining the function Ω (FO-RAW, Q-RAW) by the average value of all the errors such δn that FO-RAW = FO-RAW n and Q-RAW = Q-RAW n . Estimation device according to Claim 3, wherein the said modelling means are a ROM memory (10) or combinatorial logic comprising a first input, connected to the output of the means (2, 7, 5) for calculating an estimate of the average of the error signal ε calculated over a certain number of samples, from which estimate n FO-RAW bits are take, and a second entry, connected to the output of the means (3, 6, 8) for calculating an estimate of the average of the absolute value of the error signal ε over a certain number of samples, from which estimate bits n Q-RAW are taken. Estimation device according to claims 3 and 4, applied to a π/4-QPSK demodulator, wherein the said means (1) for calculating the error signal ε on the basis of the phase  of the output signal of the demodulator and of the ideal phase  o of this signal receive, on one input, the in-phase and quadrature components (I,Q) of the output signal of the demodulator and, on another output, the in-phase and quadrature components (I o , Q o ) of the signal having ideal phase.

Inventors: BAISSUS ERIC, BAISSUS, ERIC
Applicants: TEXAS INSTRUMENTS FRANCE, TEXAS INSTRUMENTS INC, TEXAS INSTRUMENTS INCORPORATED
Application year: 1998
Main classification: H04L27
IPC codes: H04L27