MULTICARRIER RECEIVER AND METHOD FOR USE IN DAB RECEIVER

Abstract: A digital audio broadcasting (DAB) receiver designed for receiving a signal with 1536 active carriers generated by an inverse fast Fourier transform (FFT) with 2048 points, includes the usual RF stage (14), an IF filter (20), and a demodulator (22). This is followed by an anti-aliasing filter (28), a sampler (30), and an FFT (32). The FFT is a 4096-point FFT, and is thus twice as long as required, and gathers twice as many points during each symbol period. However this is found to substantially simplify the construction of the IF filter (20) and, more particularly, the anti-aliasing filter (28).

Description: ULTICARRIER RECEIVER AND METHOD FOR USE IN DAB RECEIVER BACKGROUND OF THE INVENTION This invention relates to receivers for many-carrier signals, for example OFDM (orthogonal frequency-division multiplex) signals, such as is planneα for use in the proposed digital audio broadcasting (DA3) or digital sound broadcasting system, and also in digital television. One proposal for digital audio broadcasting for use in the United Kingdom requires seven DA3 ensembles (channels) each occupying 1-536 MHz within the overall frequency range 217-5 MHz to 230 MHz, a total bandwidth of only 12-5 MHz. The proposed spacing between ensemble centres is 1712 kHz, of which 1536 kHz is taken up by the signal, so that the spacing between the top of one ensemble and the bottom of the next is only 176 kHz. The signal bandwidth of 1536 kHz arises from the use of 1536 separate carriers spaced at spacmgs of 1 kHz. The number 1536 is chosen as three-quarters of 2048 (equals 2 to the power of eleven) . A DAB receiver must be able to receive the desired DAB ensemble to which it is tuned in the presence of a number of interfering DAB signals occupying the adjacent spectrum. Of the adjacent signals, it is the nearest neighbour which is most difficult to reject because of the high ratio of signal bandwidth to edge spacing; the edge spacing is less than one-eighth of the signal bandwidth. A typical proposed DAB receiver of this type is shown in schematic block diagram form in Figure 1 of the drawings. The receiver 100 has an antenna 112 feeding a radio frequency (RF) stage shown as an RF amplifier 114. The output of the RF amplifier is applied to a mixer 116 which receives a first local oscillator signal LOl at a terminal 113. The mixer reαuces tre frequency of the received signal, typically at around 225 MHz, to an intermediate frequency of typically aoo-.t 36 MHz. The output of the mixer 116 is applied to an IF bandpass filter 120 which passes the desired intermediate frequency in the region of 36 MHz. The output of the IF filter 120 is then applied to an I/Q demodulator circuit 122, which receives a second local oscillator signal L02 at a terminal 124. The I/Q demodulator circuit reduces the signals to baseband frequency and also separates the in-phase (I) and quadrature phase (Q) components of the signal. The output 126 of tne I/Q demodulator thus in fact comprises two signals, as indicated on Figure 1, and the subsequent circuitry is duplicated for the two signals, as is well known. The output 126 of the I/Q demodulator 122 is applied to an anti-alias bandpass filter 128, the characteristics of which are described in more detail below, and from the anti-alias filter 128 are applied to a sampler or analog-to-digital converter 130. The sampler 130 operates at 2-048 Ms/s (mega-samples per second), which is of course the same as the sample rate of the digital signals which were used to form the transmitted signal at the transmitter. In the sampler 130 the signals are now converted from analog form to digital form, and are then applied to a fast Fourier transform (FFT) circuit 132. The FFT generates a signal in the form of a sequence or series of symbol periods. The FFT has 2048 points which corresponds to the theoretical number of carriers with a sampler operating at 2-048 Ms/s, a carrier spacing of 1 kHz, and an active symbol period of 1 ms . In fact as noted above, only 1536 carriers are used, the remainder having a theoretical amplitude of zero. To achieve this places considerable demands on the filter 128. This filter should have a pass-band extending to ±768 kHz (half of 1536 kHz) but a cut-off frequency of ±1024 kHz (half of 2048 kHz) . This is a sharp cut-off and is difficult to achieve. The output 134 of the FFT is a time-based signal which is then processed using conventional receiver circuitry (not shown) . The circuit of Figure 1 will be known to those skilled in the art, and further description thereof is not necessary. Likewise, a corresponding transmitter will be known to those skilled in the art, and includes a 2048-pomt inverse FFT operating in the digital domain corresponding to the FFT 132 at the receiver. The inverse FFT receives a conventional t me-baseα signal and converts it into a many-carrier signal for transmission. Figure 2 is a spectrum αiagram showing three adjacent ensembles in the frequency spectrum. Tne numerical values are those appropriate to the DAB proposal mentioned above, and are referred to the centre frequency of the central ensemble E which is taken to be zero. One ensemble E+l is shown above this with positive values and another ensemble E-l is shown below it with negative values. The values are in kilohertz, but as the individual carriers are spaced by 1 kHz, they can equally be treated as a count of carriers. The amplitudes of the signals shown are purely arbitrary; they are shown for convenience of illustration with a slight peak at the centre of each ensemble but in theory the amplitudes should be flat. it will be seen that each ensemble extends over 1536 carriers, and that the spacing between corresponding points on the ensembles is 1712 carriers. Figure 2 also shows, for the central ensemble, the positions where the sampling frequency and the inverse appear. These fall at ±2048 carriers. The values of half the sampling frequency, fs/2, which fall at ±1024 carriers, are also shown. The value of aalf the sampling frequency is, as is well known, the M/quist limit. Frequencies which appear above half the sampling frequency can not be correctly represented by tne sampling process. These frequencies above half the sampling frequency, when subjected to sampling, give rise to spurious components in the sampled signal known as aliased components. The aliased components are commonly thought of as through the signals in the range fs/2 to fs were "reflected" about the frequency fs/2. Thus, a frequency which is g Hz below the sampling frequency fs, that is a signal of frequency (fs-g) Hz, gives rise to an alias component of frequency of g Hz. This is correct for basebands signals, but for signals above baseband, correctly what happens is that the signals above fs/2 are translated downwards by a frequency shift equal to the sampling frequency fs . Such shift occurs in fact for all integral multiples of the sampling frequency, but only the first and most powerful need be considered in practice. This is illustrated in Figure 3, which shows just the central ensemble E of Figure 2 and the ensemble E+l above t. It also shows the aliased components which arise by down-shifting the upper ensemble E+l by the sampling frequency. Those frequencies which arise in the range from above the Nyquist frequency, or half the sampling frequency, namely 1024 kHz, up to the top of the upper ensemble E+l, namely 2480 kHz, are moved downwards by 2048 kHz. The aliased components E+l now span the frequency range -1024 to +432. Of these the frequencies in the range -768 to +432 fall within the band of the wanted ensemble E. These can not be rejected by simple frequency-selective filtering. The shifted frequencies are marked in the figure by cross-hatching. The FFT circuit 132 expects only 1536 carriers out of a possible 2048, and thus inherently rejects energy in the frequency range 768 khz to 1280 kHz. Tnis upper limit equals the sampling frequency 2048 kHz minus the expected upper carrier frequency limit of 768 k.z Within the wanted oand, this includes the rejection regions marked R on Figure 3. We have thus appreciated that to cut out the interference components requires strong IF and anti-alias filtering in the filters 120 and 128 of Figure 1, in order to reject the adjacent channel energy from ensemble E+l before it reaches the analog-to-digital converter or sampler 130 To produce a sufficiently sharp cut-off may require for example a surface acoustic wave (SAW) filter for use as the filter 128. Such filters are expensive and lossy and may result in the partial loss of a number of carriers located towards the edges of the ensemble. Although the DAB transmission is very robust, nevertheless degrading the signal in this way may reduce the system margins available to combat other sources of degradation, e.g. multipath distortion or channel noise. Other types of filter may introduce less loss of signal but can introduce considerable group delay ripple into the signal. In summary, the filter requirements are quite difficult to meet without simultaneously adversely affecting the wanted signal. This problem arises independently with the IF filter 120 and with the anti-aliasing filter 128, though primarily with the latter. This problem arises in the analog processing due to the difficulty of making adequately effective filters. It might therefore be thought that the problem could be solved by constructing the filter 128 in the digital domain rather than in the analog part of the circuit. That is to say, the filter 128 (or at least the greater part of its functionality) would be placed after the sampler 130, instead of ahead of it m the signal processing chain. This would of course require the use of a higher sampling rate than 2-048 Ms/s and a rate such as 4-096 Ms/s could conveniently be chosen. Subsequent to the filter, a downsampler would be included in order to reduce the sample rate to the value of 2-048 Ms/s which can be accepted by the FFT device 132. While such an arrangement, using a digital filter, should produce theoretically improved rejection of the interfering signal, we have appreciated that it necessarily increases the signal processing requirements by a considerable amount. .It has been proposed by Muschallik, C, in "Improving an OFDM Reception using an Adaptive Nyquist Windowing", IEEE Transactions on Consumer Electronics, Vol. 42, No. 3, August 1996, pages 259-269, (see also International Patent Application W096/41458) to improve the sensitivity to frequency errors in an OFDM receiver by using a guard interval which is equal m length to the "useful" part of the symbol period, instead of being only one-quarter as long or less, as is generally considered. The receiver FFT is then of double the usual length and uses the samples in the guard interval as well as those in the useful part of the symbol period. Th s increases the carrier to noise level by 3dB. The sample rate is however unchanged. United Kingdom Patent Specification GB-A-2 , 304 , 504 and United States Patent US-A-5, 357, 502 similarly propose to increase the length of the FFT, so as to accommodate more of the received signal. * SUMMARY OF THE INVENTION The invention in its various aspects is defined in the independent claims below to wnicn reference should now be made. Advantageous features of the invention are set forth in the appendant claims . A preferred embodiment of the invention is described in more detail below, and takes the form of a DAB receiver designed for receiving a many-carrier signal with 1536 active carriers as generated by an inverse fast Fourier transform (FFT) with 2048 points. The receiver has the usual RF stage, an IF filter and a demodulator, followed by an anti-aliasing filter, a sampler, and an FFT device. The FFT is a 4096-point FFT, ana is thus twice as long as required, and gathers twice as many points during each symbol period. However, we have found that this enables the construction of the IF filter and, more particularly, the anti-aliasing filter, to be substantially simplified. BRIEF DESCRIPTION OF THE DRAWINGS The invention will be described in more detail by way of example with reference to the accompanying drawings, n which: Figure 1 (described above) is a block schematic diagram of the relevant part of a previously-proposed DAB receiver; Figure 2 (described above) is a spectrum diagram showing three adjacent ensembles of a DAB signal; Figure 3 (described above) is a spectrum diagram illustrating how the interference is generated; Figure 4 is a block schematic diagram of the relevant part of a DAB receiver embodying the present invention; and Figure 5 is a spectrum diagram illustrating how the interference is avoided in accordance with this invention. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A preferred embodiment of the invention will now be descriDed by way of example, n the form of a DAB receiver which can be used to receive tne same signal as is received by the prior art receiver of Figure 1. The receiver embodying the invention is thus completely compatible with the already-proposed transmission system in that it does not require any alterations at the transmitter. The DAB receiver 10 embodying the invention is shown m block schematic form in Figure 4. An antenna 12 feeds a radio frequency (RF) stage shown as an RF amplifier 14. The output of the RF amplifier is applied to a mixer 16, which receives a first local oscillator signal LOl at a terminal 18. The mixer 16 reduces the frequency of the received signal, typically at a frequency of around 225 MHz, to an intermediate frequency of typically about 36 MHz. The output of the mixer 16 is applied to an IF bandpass filter 20 which passes the desired intermediate frequency in the region of 36 MHz. The output of the IF filter 20 is then applied to an I/Q demodulator circuit 12, which receives a second local oscillator signal L02 at a terminal 24. The I/Q demodulator circuit reduces the signals to baseband frequency and also separates the m-phase (I) and quadrature-phase (Q) components of the signal. The output 26 of the I/Q demodulator thus in fact comprises two signals, as indicated on Figure 4, and the subsequent circuitry is duplicated for the two signals, as with the system of Figure 1. The output 26 of the I/Q demodulator 22 is applied to an anti-alias bandpass filter 28, the characteristics of which are described in more detail below, and from the anti-alias filter 28 are applied to a sampler or analog-to-digital converter 30. The sampler 30 in this case operates at 4-096 Ms/s, rather than at 2-048 Ms/s as in the system of Figure 1. In the sampler 30 the signals are converted from analog form to digital form, and are then applied to a fast Fourier transform (FFT) circuit 32. The FFT has 4096 points rather than the 2048 points of tne FFT 132 in the system of Figure 1. The output 34 of tne FFT is a time-based signal which is then processed using conventional receiver circuitry (not shown) . Thus the FFT is changed compared with the system of Figure 1, rather than trying to improve the filter 128 directly. This naturally requires a 4096 Ms/s sampler, but has the advantage that the IF filter 20 and the anti-alias filter 28 do not have to have such precise characteristics as the corresponding components in Figure 1. The IF filter can be slightly less narrow as compared to Figure 1, and thus can be of simpler construction. The anti-alias filter 28 still has a pass-band of 768 kHz, but only has to cut-off at 2048 kHz, rather than at 1024 kHz as with Figure 1. Thus although the problem has in principle nothing to do with the FFT, we have found that the problem in filter construction can be solved by making changes to the FFT which follows the filter and sampler, rather than by trying to improve the filter itself. More specifically, as w ll be described with reference to Figure 5, the longer 4096-point FFT inherently rejects adjacent channel energy in the frequency range between the wanted ensemble edge of 768 kHz and an upper limit of 3328 kHz. This is much wider than the range 768 kHz to 1280 kHz for the system of Figure 1. This increase in the inherent rejection bandwidth greatly reduces the receiver filter requirements, allowing gentler filters to be used, which causes less distortion to the wanted signal. Such filtering is also likely to give rise to lower signal loss. This has the desirable consequence that the receiver RF stage can be designeα to exmoit better linearity, since less gain is required prior to the IF stage . The rejection bandwidth arises oecaαse the FFT is still designed to extract only 1536 carriers at 1 kHz spacing; the remaining outputs (potentially containing interference) can be ignored by tne post-FFT processing. Thus, by reducing the proportion of wanted carriers to a smaller fraction of the total nurruoer of carriers, the operation has been improved. In Figure 1, the wanted carriers amount in number to three-quarters of the total number of carriers. By doubling tne total number of carriers, in Figure 4 the wanted carriers amount to three-eighths of the total number of carriers The gap between adjacent interfering ensembles thus goes up from one-quarter to five-eights of the total number of carriers . Figure 5 shows a spectrum diagram corresponding to that of Figure 2, but for the system of Figure 4 embodying the invention. The ensembles E and E+l are again shown and cover the same frequency ranges as in Figure 3. In this case, however, the sampling frequency is 4096 kHz rather than 2048 kHz, and so half the sampling frequency is 2048 kHz. Aliasing only occurs with frequencies above 2048 kHz, rather than with frequencies above 1024 kHz. This means that only a small part of the ensemble E+l, namely that lying between 2048 and 2480, is aliased. Furthermore, aliased components no longer lie within the bandwidth +768 to -768 of the wanted ensemble E. Instead, they now fall in the range -1616 to -2048, well outside the bandwidth of ensemble E. In order to provide m-band components it would be necessary for signals in ensemble E+l to extend above 3328 kHz, which it will be seen, is well above the top of the bandwidth of ensemole E-l. Thus n the embodiment described, the size of the FFT has been doubled to 4096 points, with a corresponding doubling of the sampling rate. While tnis is the most convenient value, other values could in theory be chosen, in particular other integral multiples of 2048. This oversampling increases the available processing bandwidth, to allow the IF frequency response (selectivity) to be augmented, while the time period from which the samples are drawn is unchanged. In general, the system is applicable to a transmission system which has A active carriers out of a possible P carriers, generated by use of a P-point transform. In the receiver, the transform is a Q-pomt transform, where Q is greater than P. As described Q is greater than 2A and is in fact equal to 2P. The invention has been described as applied to a DAB receiver, but it can be applied to other receivers for many-carrier transmission systems such as those produced by OFDM systems. Another example is digital terrestrial television, where part of the signal is to be transmitted in OFDM form. The numerical values here are of course different, though the principles are the same. For example, with the so-called 2k television system as currently proposed, the channel spacing is 4-46 kHz (this being the reciprocal of 224 μs) . The number of carriers used is 1705 giving a total bandwidth of 1-705 x 4-46 = 7-61 MHz. The proposed sample rate is 9-143 MHz and it is proposed to use a * 2048 point inverse FFT at the transmitter and a 2048 point FFT at the receiver. In accordance with this invention as applied to an embodiment for reception of such a television signal, the sample rate is preferably raised, in fact doubled, to 13-286 MHz, and a 4096-point FFT is again used. Similar principles are applicable to an 8k television system. While the invention has been descr oed as using Fourier transforms to produce and process the many-carrier signal, other transforms could in principle be employed. As is seen from tne above description, the problem of making the filters in a many-carrier oroadcast receiver is overcome not by improving the filter construction, nor by putting the filter in the digital domain, but rather by changing the size of the FFT which is used to one which nas greater redundancy. This then enables the constraints on the construction of the filters to be very substantially reduced.

Claims: CLAIMS 1. A many-carrier transmission system, comprising a transmitter which includes inverse transform means for generating from an input signal a many-carrier transmission signal having A active carriers out of a possible P carriers (A<P) by use of a ?-poιnt transform, the transmission signal consisting of a sequence of symbol periods, and a receiver for receiving tr.e many-carrier signal transmitted by the transmitter, the receiver including: means for band-pass filtering and for sampling the received many-carrier signal at a predetermined sample frequency chosen to gather a multiple of P points during one symbol period; and transform means for receiving the filtered and sampled many-carrier signal and for generating a time-based signal therefrom; in which the transform means at the receiver operates with a Q-point transform where Q is greater than P. 2. A many-carrier transmission system according to claim 1, in which Q is greater than twice A. 3. A many-carrier transmission system according to claim 1, in which Q is equal to or greater than twice P. 4. A many-carrier transmission system according to claim 1, in which P is 2048 and Q is a multiple of 2048. 5. A many-carrier transmission system according to claim 1, in which P is 2048 and Q is 4096. 6. A many-carrier transmission system according to any of claims 1 to 5, in whicn the Dand-pass filtering and sampling means includes for the band-pass filtering an anti-aliasing filter coupled to receive tne received signal, and for the sampling a sampler coupled to receive the output of the anti-aliasing filter and supplying an output to the transform means. 7. A many-carrier transmission system according to any of claims 1 to 5, in which the transform means comprises a Fourier transform device. 8. A method of transmitting a many-carrier transmission signal, comprising the steps of generating from an input signal a many-carrier transmission signal having A active carriers out of a possible P carriers (A<P) by use of a P-pomt transform, the transmission signal consisting of a sequence of symbol periods, and transmitting the resultant many-carrier signal, and receiving the many-carrier signal transmitted by the transmitter, the receiving step including band-pass filtering and sampling the received many-carrier signal at a predetermined sample frequency chosen to gather a multiple of P points during one symbol period, and generating by use of a transform a time-based signal from the filtered and sampled many-carrier signal; n which the transform at the receiver operates with a Q-point transform where Q is greater than P. 9. A method according to claim 8, in which Q is greater than twice A. , 10. A method according to claim 8, in which Q is equal to or greater than twice P. 11. A method according to claim 8, in wrich P is 2043 and Q is a multiple of 2048. 12. A method according to claim 8, --^ iich P is 2043 and Q is 4096. 13. A method according to any of claims 8 to 12, in which the band-pass filtering and sampling step includes for the band-pass filtering the step of antι-al_as filtering the received signal, and for the sampling step sampling the anti-alias filtered signal to supply an output to the transform. 14. A method according to any of clams 8 to 13, in which the transform comprises a Fourier transform. 15. A receiver for receiving a many-carrier signal consisting of a sequence of symbol periods, the receiver comprising: means for band-pass filtering and for sampling a received many-carrier transmitted signal which includes a predetermined number A of active wanted carriers at a predetermined sample frequency chosen to gather a multiple of P points during one symbol period; and transform means for receiving the filtered and sampled many-carrier signal and for generating a time-based signal therefrom; and in which the transform means at the receiver operates with a Q-pomt transform where Q s greater than twice A. 16. A receiver according to claim 15, in which Q is 4096. 17. A receiver according to claim 15 or 16, in which the transform means comprises Fourier transform means.

Inventors: MOSS PETER NEIL, MOSS, PETER, NEIL
Applicants: BRITISH BROADCASTING CORP, MOSS PETER NEIL, BRITISH BROADCASTING CORPORATION, MOSS, PETER, NEIL
Application year: 1998
Main classification: H04L27
IPC codes: H04H20, H04H40, H04L27