HIGH-EFFICIENCY ELECTRIC MOTOR OF ELECTRONIC COMMUTATION TYPE

Abstract: A high-efficiency electric motor of electronic commutation type having a single stator unit and a single rotor unit comprises a first electrical submachine (M1) and a second electrical submachine (M2), in which the first submachine (M1) is fed by a voltage source (Vb) and is associated with a sensor (Rfb) for measuring the current absorbed from said feed. The first submachine (M1) comprises at least two windings characterised by an inductance (Lf1), a resistance (Rf1), and induced electromotive force (Ef1) and a switch (P) connected in series. The second electrical machine (M2) is fed by a capacitor (C), which is charged at a controlled voltage (Vc). For each of said first windings there is provided a diode (D), having one of its poles connected to the end of the respective winding, which is connected to the switch (P), and the remaining pole connected to one of the ends of the capacitor (C) thus charged at a controlled voltage (Vc). Moreover the first submachine (M1) is pulse-modulation driven and performs the function of power supply for the submachine (M2) by charging the capacitor (C) at the voltage (Vc) via the diodes (D).

Description: HIGH-EFFICIENCY ELECTRIC MOTOR OF ELECTRONIC COMMUTATION TYPE DESCRIPTION Field of the invention This invention relates to a high-efficiency electric motor of electronic commutation type. Background of the Invention High-efficiency electrical machines of electronic commutation type, hereinafter known as ECMs, operate with pulse modulation and generally at ultrasonic frequencies, with absorption of very high ripple current pulses. Without the use of a costly and bulky L-C filter in the feed line, the conducted and radiated electrical disturbance levels would be greater than allowed by current regulations. To reduce costs, said filter can be replaced by one of active type able to decouple the current absorbed by the electric motor from the battery current. A known and particularly effective implementation, in terms both of cost and performance, is to interpose between the battery and the ECM a step-up converter current-controlled by means of R re on the basis of control information C re originating from the ECM, which is compared with a velocity input V, βt by known methods. Said converter is characterised by operating with V c greater than V and by absorbing from the battery an essentially continuous current (of constant delivered power) with a ripple as small as desired, achieved by dimensioning the inductor L and the switching frequency by known methods. The waveform of the current i b is shown in FIG. 2, which shows the typical times associated with the operation: 1/T is the switching frequency, T on and T of£ are the on and off times of the electronic power switch P (FIG. 1); also shown are the ripple superimposed on the mean absorbed current and the composition of i b , consisting of the sum of i p and i D , this latter being integrated by the capacitor C to provide i 2 (from an essentially continuous V c ) which powers the ECM. Summary of the Invention The object of the invention is to achieve the operability of the schematic of FIG. 1 essentially in terms of the waveform of the current absorbed from the battery, while significantly reducing cost and bulk by eliminating the inductance L and the switch P. As it is not possible to eliminate these components from an operational viewpoint, the invention proposes a solution which utilises certain switches and certain windings of the ECM, already present for its normal operation, to also perform the function of switch P and inductance L. This object is attained according to the invention by a high-efficiency electric motor of electronic commutation type, comprising a single stator unit and a single rotor unit, characterised by comprising a first electrical submachine and a second electrical submachine, in which: said first submachine is fed by a voltage source and is associated with a sensor for measuring the current absorbed from said feed; said first submachine comprising at least two windings characterised by an inductance, a resistance, an induced electromotive force and a switch connected in series; - said second electrical machine is fed by a capacitor, which is charged at a controlled voltage; for each of said first windings there is provided a diode, having one of its poles connected to the end of the respective winding, which is connected to the switch, and the remaining pole connected to one of the ends of the capacitor thus charged at a controlled voltage; the first submachine is pulse-modulation driven and performs the function of power supply for the submachine by charging the capacitor at the voltage via said diodes. Brief description of the drawings The machine according to the invention is described hereinafter with reference to the accompanying drawings, in which: FIG. 1 is the electrical schematic of an electric motor of known type; FIG. 2 shows the waveforms of the current through the motor of FIG. 1; FIG. 3 is the basic electrical schematic of the electric motor according to the invention; FIGS. 4 and 5 are two electromagnetic structures forming the electric motor of the invention; FIG. 6 is the specific schematic of the electric motor of the invention; FIG. 7 shows the phase emfs of the motor of the invention; FIG. 8 is a simplified schematic corresponding to that of FIG. 6; FIG. 9 is a further electrical schematic of the machine of the invention; FIG. 10 is a waveform diagram; FIG. 11 is, the electrical schematic of the motor of the invention provided with protection devices; - ~ FIGS. 12 and 13 are further waveform diagrams; and FIG. 14 is the electrical schematic of an additional circuit for the electric motor of this invention. Detailed description of preferred embodiments Essentially according to the invention, the inductance L and switch P of FIG. 1 are integrated into a suitably structured ECM, controlled and dimensioned to add to its electric motor function the function of active filter, so covering by itself the overall operability of the schematic of FIG. 1. The first feature of the ECM proposed by the invention (FIG. 3) is that it operates as two submachines which mechanically combine their contributions at the same rotor of the ECM whereas electrically they operate and are controlled as two separate machines. The first, known hereinafter as Ml, is powered by the battery at voltage V b , whereas the second, known hereinafter as M2, is powered by a capacitor C charged to a voltage C c by the operation of Ml as described hereinafter. The scheme is completed by the fast diodes D connected to the capacitor C as in FIG. 3. The velocity input V 8βt and the signals of the Hall position sensors are also shown. The second feature is that in order to also perform the function of the inductance L and the switch P of FIG. 1, the submachine Ml must be designed with a unipolar structure with two or more windings (depending on the number of phases to be determined and the number of windings to be powered in parallel) with the magnetic coupling between them as loose as possible. The inductances of its windings and the switches P already proposed for their normal PWM driving provide the L and P functions of FIG. 1. The third feature is that the submachine M2 can have a different number of phases and windings than the submachine Ml, with any magnetic coupling between them, but magnetically decoupled from the windings of Ml. The fourth feature is that the driver of M2 is totally independent of that of Ml. It can therefore be of unipolar, bridging, linear or PWM type and is characterised by having a control function (for example a control feed-back on V c ) which ensures that under all operating conditions the current induced by the operation of Ml via the diodes D is totally absorbed by M2. Without limiting the generality of the aforedescribed principle of operation, for greater clarification and for providing the main design principles, reference will be made to a two-phase battery powered unipolar brushless motor of permanent magnet type. Two electromagnetic structures which implement the aforesaid magnetic coupling conditions are shown in FIGS. 4 and 5 by way of non-limiting example. In particular, for the same nominal ECM operating conditions and the same number of poles, the structure of FIG. 5 has a lower phase inductance and a lesser demagnetising reaction (1/3 of that of the structure of FIG. 4). The specific schematic which achieves the said principles (FIG. 1 and FIG. 3) is shown in FIG. 6. To complete the control electronics, in addition to that already described it includes two signals V^ for operating by known circuits (clamping circuits) a protection at overvoltages exceeding the VDSS allowed by the switches P2. These latter together with other circuit details are known and do not form part of the inventive idea, and will therefore not be referred to hereinafter. The chosen two-phase structure is for example of known type with four unipolar windings powered as two single-phase machines (at full haIf-wave) . The first single-phase machine (consisting of PHASE 1 and PHASE 3) covers the role of the submachine N2 and is powered at V c . The emfs of each phase (e F1 , e F2 , e F3 , e p4 ) are shown in FIG. 7, where it can be seen that they are out of phase by 90 electrical degrees. The magnetic structure, the seat of the magnetic flux generated by the currents in each winding of the submachine Ml (identified in FIGS. 4, 5 and 6 as PHASE 1 and PHASE 3) , must be such as to ensure that the inductances of these windings are as mutually decoupled as possible to prevent absorbed current gaps during switching between one winding and the next in the driving sequence (a known problem when mutual inductance exists between the two) and that the inductive couplings with the windings of M2 are marginal. This is achieved by the presence of non-wound decoupler teeth (indicated by T d ) and winding the two phases (PHASE 1 and PHASE 3) on physically separate teeth (see FIGS. 4 and 5). Said M2 windings also operate as an electric motor generating an active torque, as they suitably engage the pertinent emf half-wave by known methods (e.g. suitable decoding of Hall position sensors). The magnetic structure, the seat of flux generated by the currents in each winding of the submachine M2 (identified in FIGS. 4, 5 and 6 as PHASE 2 and PHASE 4), must ensure in this case a very tight magnetic coupling between them to enable the stored magnetic energy (from the windings which cease to conduct to those which begin to conduct) to be transferred during switching with minimum losses via the diodes D2 (known operation) . This is achieved by winding said phases on the same teeth (see FIGS. 4 and 5). As the two submachines operate in parallel in providing the desired mechanical power it is generally advantageous to dimension them such that, at least under nominal conditions, both the mechanical power supplied and the losses are divided into equal parts. The design data for said operating point (n) are: p β ch ( n ) mechanical power RPM(n) velocity η(n) efficiency V b feed voltage Knowing the design data, the geometry and the materials chosen for constructing the machine, the iron, ventilation and friction losses P fβ,v , a ( n ) can be predicted by known methods. The value of R ra is chosen such that the voltage drop across it can be considered negligible as a first approximation, so that to simplify the calculations the diode is simulated as an ideal diode with a resistor equal to R p i in series (see FIG. 8). By way of example, for the machine of FIG. 8 the equivalent scheme shown in FIG. 6 can be used, which shows the essential components for dimensioning the two machines, these being: L £1 inductance of each winding of Ml - - R fl resistance of each winding of Ml E fl mean emf per half wave at the nominal velocity of each winding of Ml R P1 internal resistance of the power switch (e.g. MOSFET) for each winding of Ml. FIG. 8 also shows the corresponding elements for M2. The two submachines (Ml) and (M2) must be designed as follows. Dimensioning of submachine Ml: The first element immediately obtainable is i 1{n) from η(n) = Paβohtn) V b(n)il(n) hence i 1(n) = 'P mmch iI . ) / V b(n) n (n) (eq. 1) Equation 1, together with cost considerations and other known operational aspects of the switch PI, enables its type to be identified and hence R P1 to be qualified as an item of data. Having identified i 1(n) and R pl , E fl (n), E fK iooo) a *ιd R f i( n ) can be obtained. From the known relationship P w = P^ ch + P fβ , V(a = El and remembering that the power has to be distributed equally between the machines Ml and M2, the for Ml: E fl (r) i 1 (n) = [V b{n) i 1 {n) η (n) + P fβ ,v, a (n) ] /2 hence E fl {n) = J [V b{n) η ( n) + P £β , v ,a( n > / ii (n) l which by replacing i 1 (n) by eq . 1 gives : E f i (n) = έ V b (n) η ( n) [ l + P £β , v ,a<n> / *m.c (n) 3 ( ®q . 2a) As P fβ V , a(n) is negligible compared with Pn»c (n) / ( β q. 2a) can 5 be rewritten as E.ι(n) * £ V b(n) η(n) (eq. 2b) from which E fl(10 oo ) can be obtained as follows: E £ ι(1000) - [E fl(n) / RPM(n)] • 1000 (eq. 3) Hence using known formulas the number of turns of the winding and the value of L fl can be calculated. To obtain R f i (n i ar * energy balance can be used in which the machine Ml absorbs 50% of the total power. Hence: [Efi(n) + (Rfl + R Pl)il(n) l il(n) = έ V b(n> il(n) giving E £1 (n) + (R fl + R P1 ) i 1(n) = J V b(B) from which R fl = [V b(n) /2 - E fl(n) ]/i 1(n) - R P1 (eq. 4) Dimensioning of submachine M2: Defining T on and T off as the on and off times of the switches PI respectively, - T on + T off , D = T on /T, T off /T = (1 - D). Regardless of the voltage V c across the capacitor C, its charging current can be obtained from the always valid relationship: i 2 - ii T of£ /T = ii(l - D) which at the nominal operating point can be written as i 2 (n) = iκ n )(l ~ D(n)) (eq. 5) The relationships between M2 and Ml for their respective characterising elements can now be obtained. Remembering the condition of equal power, then: E f2(n) 'i2(n) = E f 1(r) .i 1(n) hence E f2(n) = E fl(n) . i 1(n) / i 2 (n) and finally E £2(n ) = E £1(n) /(1 - D (n) ) (eq. 6) Remembering also the condition of equal dissipated power, then: hence and finally (eq. 7) The only unknown is D (n) , which can be obtained from from which, having assumed RD = Rpl (V b " [E fl + (R fl + / L fl T = (Rpiii + V c - [V b - (E fl + R fl ii)]) T o£f /L fl T (eq. 8) Putting A ■ V b - [E £1 + (R £1 + R P ι)iι], then: D - (V e - A)/V c (eq. 9) Hence, remembering (eq. 4), 1 - D (n) = V b(n) . V β(a) /2 (eq. 10) from which it can be seen that having fixed V b , (1 - D (n) ) is defined unambiguously by V c(n) . The three ensuing conditions help to define V e(n) unambiguously. These are: Condition 1 In order for current not to circulate through that winding of the submachine Ml which with its emf, the sum of the motional part E fl(n) and the transformer part E^^* due to undesirable coupling between the windings of the submachine Ml and between these and those of the submachine M2, would give a negative contribution to the development of mechanical power, the voltage V Dsl{o£f) across the power switch P 1(βf ) connected to said winding must be less than the voltage across the capacitor C. Only in this manner can the diode D 1(o££) be polarised inversely and hence current cannot pass therethrough. The following condition must therefore be satisfied (see FIGS. 8 and 9) : V c < » ) ≥ os Koff) - V b(n) + E fl(n) + E n u a ) (cond. 1) Condition 2 As the maximum voltage V DS2(0ff) across the power switch P2 occurs during the time interval in which that winding of the submachine M2 connected to it is inactive, then: DS2(o£f) = V 0(auϋc) + E f2(πuuc) = 2 V c(πuut) ; hence in order for the rupture voltage V DSS2 of the power - I - switch P2 not to be exceeded, the following condition must be satisfied: 2 V c(IBax) < V DSS2 (cond. 2) Condition 3 Remembering that: - the coupling between the windings of the submachine M2 must, as stated, be as high as possible; - the transfer of magnetic energy, which occurs through D2 during switching between the windings of submachine M2, is less dissipative the higher the difference between the feed voltage, which in this case is V c , and the transient overvoltage V ts2(o£f,t) (made as close as possible to V DSS2 by said clamping circuits) which appears across the power switch P2 when it opens; - the cost of the capacitor C increases with its rated voltage; it is apparent that V c(n) must be as low as possible (cond. 3). Given that in practice: E £ i {n) + jnKnj * £ V b (n) then (see (cond. 1)) V e(a) * 3/2 V b(B) (eq. 11). From (eq. 10) and (eq. 11) the following are also obtained: i 2(n) = iι (n) /3 (eq. 12.1) Equations 12.1 - 12.4, which unambiguously determine the dimensioning of the submachine M2, show an interesting aspect from the constructional viewpoint, namely that for the two submachines, wire of the same cross-section can be used, with a different number of wires in parallel for the two submachines. If l m is the mean turn length identical for all windings of the two submachines, S cl the wire cross-section of each winding of the submachine Ml and S c2 the wire cross- section of each winding of the submachine M2, then: Rfi - P (1« N βl )/ C cl (eq. 13.1) R f2 - P (1» N B2 )/ C c2 (eq. 13.2) Given that from (eq. 12.2) it can be deduced that the number of turns N βl of each winding of the submachine Ml must be 1/3 the number N s2 of each winding of the submachine M2: N βl = 1/3 • N s2 (eq. 13.3) From (eq. 12.3) and (eq. 13.1-13.3): p(l m • N 82 )/S c2 = 3 2 p (1. -N βl )/S cl = 3 2 p(l m • N β2 /3)/S cl hence S o i = 3S c2 (eq. 14) This latter shows that the winding of the submachine Ml can be formed by positioning in parallel three wires of cross-section identical to that of the single wire used for the winding of the submachine Ml. A PWM control strategy at fixed frequency is normally implemented on step-up converters of the type shown in FIG. i. Given that, as clarified in the description of the inventive idea, the function of the inductor L of FIG. 1 is performed by windings which are the seat of induced emf, a strategy such as tY ~ aforegoing would make it difficult to contain the attery current ripple within predetermined limits. For this reason the control strategy adopted is of hysteresis type, which acts only on the on phase of the submachine Ml and, in accordance with known methods, maintains the current is absorbed by the ECM, as measured through the resistor R ra , within predetermined maximum and minimum values such as to make the ripple as small as desired compatible with the technical limitations related to the state of the art of the switching devices used. This naturally means that the switching frequency of the power switches of the submachine Ml is not set but is directly related to its electrical parameters (inductance, emf, feed voltage). Conveniently, a control strategy is used for the voltage V c across the capacitor C which for each delivered torque and rotational velocity condition satisfies the said (cond. 1), while maintaining the difference between V c and V Ds κ o f ) as small as desired by known methods. The said strategy enables the battery current to be fully - controlled during switching between windings of the submachine Ml. If during switching between windings of the submachine Ml it happens that the current in the phase which is switched off decreases more rapidly than the current increase in the phase which is switched on, the current is fails to below the minimum set value. If in contrast when one phase is switched off the current decreases more slowly than the current increase in the phase which is switched on, the is control maintains it within the preset limits. To obtain this condition it is necessary that during the switching time the average value of E £1 , known as E £1#!4v is such that V b " E tltΛVq > V 0 - (V b - E fl#w ) As V c « 3/2 V b , necessarily E ιavg < 0.25 V b . Given that this is achieved by simply anticipating switching (already necessary for operation of the submachine M2 and easily implemented), the absorbed current ripple is hence easily controllable in any event. A filter for eliminating conducted and radiated electrical disturbances is conveniently positioned in the ECM feed line (see FIG. 11) and is of much smaller cost and size than that required for an ECM which does not implement the inventive idea. The simplest way of protecting a battery-powered ECM is to connect a power diode in series with the operating relay. Besides being costly and bulky, this diode introduces a voltage drop (typically 0.7 Volt) and hence reduces the EM efficiency (for equal absorbed power). The operating relay, which is key-operated, has to withstand a switch-on current which is so high as to require : unacceptable overdimensioning. According to the schematic shown in FIG. 11 the ECM is instead directly powered by the battery via the relay RL controlled by the electronic control unit ECU. A lower- power diode D p and a ballast resistor R z are connected as shown in FIG. 11. Given that the electronic control unit which controls the relay RL is key-powered via D p , the ECM is protected against polarity inversion. The ballast resistor R,. prolongs the duration of the current pulse which charges the capacitors C and C F when the starting switch is operated, so limiting the extent of the dV/dt to which the capacitors are subjected and preventing passage of destructive current through the switch. The electronic control unit ECU measures the voltage across the resistor R j . and enables the relay RL only when this voltage, and hence the switch-on current, fails below a predetermined safety level. Referring to eq. (11) V c * 3/2 V b# there are some cases (for instance to lower the rms current through the capacitor C, to lower the current through the switches of submachine M2, etc. ) in which it is necessary to have V c > 3/2 V b . In that case it could happen that during the commutation between the phases of the submachine Ml, the current in the phase which is switched off decreases more rapidly than the current increase in the phase which is switched on: the battery current will fall out of the prescribed tolerance-band. To avoid the fall of the battery current it is necessary to add an electronic circuit (FIG. 14) to control the current in the phase which is switched off. This is attained, as described below, by artificially prolonging the conduction interval of each phase of submachine Ml, feeding to the gate of the corresponding MOSFET a clock signal logically anded with the pwm signal that normally controls the phases of submachine Ml in order to maintain the battery current within the prescribed tollerance- band. The decrease of the phase current vs time (slope) is controlled at a value such to avoid battery current to fall out of the above mentioned. The logic keeps the MOSFET definitively off when the phase current reaches zero. The behaviour of the circuit will be explained for one of the two phases (named 1) of submachine Ml, providing that complementary circuitry is used for the other(s). Referring to FIGS. 12 and 13, let the phase 1 switched off. V D1 = voltage at the drain of MFT1 V c = voltage across the capacitor C clock = square wave with duty-cycle value less than 50% the duty-cycle of the pwm signal and frequency value at least greater than three times the frequency of pwm signal hall - the Hall effect sensor signal which switches on phase 1 pwm = signal which normally controls the phases of submachine Ml in order to maintain the battery current within the prescribed tolerance-band. When HALL goes down to <low>, MFTl is switched (momentary) off, V D1 becomes greater than V c , b 2 goes to <high>, y α goes to <high> and q : will latch clock, out^cloc^MFTl will be controlled by pwm anded with the clock one (see FIG. 12). When the current through the phase 1 reaches zero and q λ (latched to clock) switches off MFTl, V D1 cannot override V C( B x goes to <low> and when clock goes to <low>, y x goes to <low>; suddenly q j will go to <low>, out x will go down to <low> and MFTl will be definitively switched off (see FIG. 13).

Claims: - - CLAIMS 1. A high-efficiency electric motor of electronic commutation type, comprising a single stator unit and a single rotor unit, characterised by comprising a first electrical submachine (Ml) and a second electrical submachine (M2), in which: said first submachine (Ml) is fed by a voltage source (V ) and is associated with a sensor (R ra ) for measuring the current absorbed from said feed; said first submachine (Ml) comprising at least two windings characterised by an inductance (L fl ), a resistance (R fl ), an induced electromotive force (E ) and a switch (P) connected in series; said second electrical machine (M2) is fed by a capacitor (C), which is charged at a controlled voltage (Vc); for each of said first windings there is provided a diode (D), having one of its poles connected to the end of the respective winding, which is connected to the switch (P), and the remaining pole connected to one of the ends of the capacitor (C) thus charged at a controlled voltage (V c ) ; the first submachine (Ml) is pulse-modulation driven and performs the function of power supply for the submachine (M2) by charging the capacitor (C) at the voltage (V c ) via the diodes (D) . 2. A motor as claimed in claim 1, characterised in that said second submachine (M2) comprises at least two windings, characterised by an inductance (L f2 ), a resistance (R f2 ), an induced electromotive force (E f2 ) and a switch (P2) connected in series. 3. A motor as claimed in claims 1 and 2, characterised in that the windings of each submachine (Ml, M2) are connected in parallel. 4. A motor as claimed in claim 1 or 2, characterised by comprising a diode (D2) connected in parallel with said switch (PI, P2). 5. A motor as claimed in claim 2, characterised by comprising an electronic control unit (ECU) arranged to control a plurality of power switches (P) on the basis of the following signals: controlled voltage (V c ) of the capacitor (C), current circulating within the voltage source (V b ) , voltage across the switches (P2) of the second submachine (M2), cinematic quantities of the motor (V, βt , Hall). 6. A motor as claimed in claim 1, characterised in that said inductors (L fl ) are electromagnetically decoupled from each other, or the windings of said inductors (L fl ) are inductively separated. 7. A motor as claimed in claim 2, characterised in that the number of inductors (L f2 ) of said second submachine (M2) is equal to the number of inductors (L fl ) of the first submachine (Ml) . 8. A motor as claimed in claim 2, characterised in that the number of inductors (L £2 ) of said second submachine (M2) is different from the number of inductors (L £1 ) of the first submachine (Ml). 9. A motor as claimed in claim 2, characterised in that said inductors (L £2 ) of the second submachine (M2) are electromagnetically coupled together. 10. A motor as claimed in claim 1 or 2, characterised in that said inductors (L £2 ) of the second submachine (M2) are electromagnetically decoupled from said inductors (L £1 ) of the first submachine (Ml). 11. A motor as claimed in claim 1, characterised in that the driver of the second submachine (M2) is independent of the driver of the first submachine (M2). 12. A motor as claimed in claim 11, characterised in that the driver of the second submachine (M2) is chosen as one of the following types: unipolar, bridging, linear, PWM. 13. A motor as claimed in claim 6 or 10, characterised in that between two adjacent stator teeth provided with windings, there is interposed a ferromagnetic element (T d ) without windings. 14. A motor as claimed in claim 13, characterised in that the windings of the first submachine (Ml) are formed such that the respective phases are wound on different teeth separated by a ferromagnetic element (T d ) without windings. 15. A motor as claimed in claim 13, characterised in that the windings of the second submachine (M2) are formed such that the respective phases are wound on the same teeth. 16. A motor as claimed in claim 6 or 10, characterised in that each stator tooth provided with windings is split into a fork, between the ends of which there is interposed a ferromagnetic element (T d ) without windings. 17. A motor as claimed in claim 16, characterised in that the windings of the first submachine (Ml) are formed such that the respective phases are wound on the forked ends of two adjacent teeth, said ends being separated by said ferromagnetic element (T d ) without windings. 18. A motor as claimed in claim 1, characterised in that the power source is chosen as one of the following: d.c. voltage generator, storage battery, rectified a.c. voltage generator. 19. A motor as claimed in claim 11, characterised in that said driver is of current hysteresis type, said driver acting on the windings of said first submachine (Ml) to maintain the current circulating through said voltage source (V b ) within a required range, 20. A motor as claimed in claim 19, characterised in that the switching frequency of the switches (P) of said first submachine (Ml) depends on the electrical parameters of said first submachine (Ml) . 21. A motor as claimed in claim 1 or 5, characterised in that the first submachine (Ml) is powered by the voltage source (V b ) via an L-C filter designed to eliminate the conducted and radiated disturbances, and at least one relay (RL) which is controlled by the electronic control unit (ECU) and is operated by a key circuit, a power diode (D p ) and a resistor (R-.) being connected in series and overall positioned in parallel to the electrical branch intercepted by the relay, said electronic control unit (ECU) acting on the basis of the voltage across said resistor (RJ to feed a signal by the operation of said relay (RL) at the moment in which the voltage across said resistor (R z ) falls below a predetermined value. 22. A motor as claimed in claim 1 or 21, characterised in that if the value of the controlled voltage (V c ) at which said capacitor (C) is charged is equal to 1.5 times the value of the voltage across said voltage source (V b ) , the windings of the inductors (L £1 ) of said first submachine (Ml) are formed by positioning in parallel a number of wires equal to three times the number of wires used for the windings of the inductors (L £2 ) of said second submachine (M2) and having the same cross-section. 23. A motor as claimed in claim 1, characterised in that said motor is a unipolar two-phase electrical machine with four inductor windings, in which two of said windings are inductively separated from each other to form the first submachine (Ml) and the other two are inductively coupled together to form the second submachine (M2), the driver of said second submachine (M2) acting via at least two MOS transistors so as to utilised the parasite diodes of said transistors for energy recovery. 24. A motor as claimed in claim 5, characterised in that the electronic control unit (ECU) provides control of the voltage (V c ) across the capacitor (C) such that: V c > V b + E l + Em where V is the feed voltage; E fl is the end per half wave of each of the windings of the submachine (Ml); Em is the induced emf due to undesirable coupling between the windings of the submachine (Ml). 25. A motor as claimed in claim 1 or 21 characterised in that if the value of the controlled voltage (V c ) at which said capacitor (C) is charged is more than 1.5 times the value of the voltage across said voltage source (V ) , an electronic circuit is provided to -11- control the current in the phase being switched off. 26. A motor as claimed in claim 25 characterised in that said electronic circuit provides a pwm signal fed to the gate of the corresponding MOSFET to artificially prolong the conduction interval of each phase of submachine Ml. 27. A motor as claimed in claims 25 and 26 characterised in that an electronic logic is provided to definitively switch off said pwm signal once the current in said phase reaches the zero value.

Inventors: DE FILIPPIS PIETRO, DE FILIPPIS, PIETRO
Applicants: BITRON SPA, FILIPPIS PIETRO DE, BITRON S.P.A., DE FILIPPIS, PIETRO
Application year: 1997
Main classification: H02P6
IPC codes: H02K3, H02K16, H02K29, H02M3, H02P6