A double forward converter with soft-PWM switching

Abstract: A double forward DC-to-DC converter with soft-PWM switching provides two voltage pulses to the output low-pass filter within one switching cycle (hence, the name double forward). The double forward DC to DC converter includes a transformer having a single primary winding and a two secondary windings. A main switch, under the control of a PWM control circuit, is connected is series with the primary winding. An auxiliary switch is coupled across with the main switch by way of a resonant capacitor. The resonant capacitor and auxiliary switch are used to automatically reset the transformer core while the main switch is on back to the voltage source connected to the transformer primary. A diode and snubber capacitor is connected across the drain-source terminals for both the main and auxiliary switches for minimizing switching losses. The current mirror used in prior art is abandoned in order to avoid the locked-up mode of operation. The secondary windings of the transformer are connected to a pair of saturable reactors which, in turn, are connected to rectifier diodes in order to eliminate the reverse recovery losses of the rectifier diodes. A freewheeling diode is connected across the secondary winding and cooperates with the saturable reactors to maintain the load current to the secondary windings during the transitions of the main and auxiliary switches.

Description: 1. FIELD OF THE INVENTION The present invention relates to a DC-to-DC converter circuit, and more particularly to a double forward DC-to-DC converter circuit with soft-PWM switching which includes a main switch, an auxiliary switch, rectifier diodes and a transformer for transferring energy from a primary winding of the transformer to a secondary winding when the main switch is under the control of a PWM circuit. The DC-to-DC converter circuit includes circuitry for automatically resetting the energy stored in the transformer winding when the main switch is turned off, without the possibility of being locked up as well as circuitry for soft switching of the rectifier diodes to eliminate reverse recovery losses associated with the rectifier diodes connected to the transformer secondary winding. The main and auxiliary switches are both turned on at zero voltage while snubber capacitors are provided to minimize turn-off losses. 2. DESCRIPTION OF THE PRIOR ART DC-to-DC converters are used to convert an unregulated source of DC power into a source of constant voltage for use in various applications. Such DC converters normally include a transformer having primary and secondary windings. A switch, for example, a solid state switch, is connected to the primary winding in order to control energy transfer from the primary to secondary winding. In PWM-controlled converters, the switch is normally under the control of a pulse width modulator (PWM) circuit which varies the duty cycle, defined as switch on time over switching period. With the increase of switching frequency to reduce the size and weight, as constantly required in aerospace and military applications, the switching losses increase rapidly. To cope with this, DC-to-DC power converter designers employ various schemes to eliminate or to minimize losses associated with switching at solid state switches. These schemes are generally referred to as various techniques of soft switching. Among all kinds of soft switching schemes, the most promising ones are those which have zero-current or zero voltage switching at turn on and turn-off transitions, while keeping voltage and current stresses similar those in a PWM (hard-switched) converter. This is so because stress levels for switch voltage and current similar to those in a PWM (hard-switched) converters represent the best possible efficiency in power transfer. This group of soft switching can be called soft-PWM switching. In recent developments the popular phase-shift scheme for soft switching comes close to realize soft-PWM waveforms. However, its applications are limited to double-ended converters, such as half-bridge and full-bridge converters. Most recent developments of soft switching schemes for single-ended converters, such as forward and flyback, utilize a mechanism known as the current mirror to reset the transformer automatically. Examples of such circuitry are disclosed in U.S. Patent Nos.: 4,441,146; 4,809,148; 4,959,764 and 5,126,931. Examples of such circuitry is also disclosed in the following publications: Constant Frequency ZVS Converter with Integrated Magnetics , by J.A. Bassett, publication no. 0-7803-0485-3/92, 1992, IEEE, pp. 709-716; High Frequency, Soft Transitions Converter , by I.D. Jitaru, IEEE publications no. 0-7803-0982-0/93, 1993, IEEE, pp. 880-887; Switched Snubber for High Frequency Switching , by K. Harada and H. Sakamoto, IEEE publication no. CH2873-8/90/0000-0181, 1990, IEEE, pp. 181-187 and Design Techniques for Transformer Active Resets Circuits at High Frequencies and Power Levels , by B. Carsten, HFPC May 1990 proceedings, pp. 235-245. Such circuits typically include a second solid state switch and a capacitor for transferring the magnetizing current stored in the transformer winding when the main solid state switch is opened, recycling energy back to the DC voltage source connected to the transformer primary winding. Such circuitry causes the main and auxiliary switches to be turned on at zero voltage and includes a lossless snubber, normally a capacitor connected in parallel across the switch. The snubber is used to minimize the voltage stress and losses when the switch is turned off. In addition, such circuits are configured to discharge the snubber capacitor prior to closing of the switch to eliminate the turn on loss and to minimize the voltage stress across the switch when the switch is turned on. Examples of such circuitry are disclosed in U.S. Patent Nos. 4,959,764, 5,126,931 and 5,231,563. Note that the technique of the current mirror, used in the cited patents, may cause the circuit to operate in a mode locked-up to a subharmonic frequency. The locked-up mode of operation forces the output voltage to go out of regulation and may potentially damage the switching elements. An additional protection circuit is used to guarantee safe operation of the converter. As disclosed in U.S. Patent Nos: 4,809,148; 4,441,146; 4,959,764 and 5,126,931, DC-to-DC converters normally include one or more rectifier diodes, connected to the transformer secondary winding. Typically one rectifier diode is connected in parallel across the secondary winding of the transformer while a second rectifier diode is connected in series with the load. In any switching converters, the rectifier diodes are subject to what is known as reverse recovery losses which result when the biasing across the diode quickly reverses. Various techniques are known for minimizing the reverse recovery losses of the rectifier diodes, for example, as disclosed in High Frequency, Soft Transitions Converter , by I.D. Jitaru, publication no. 0-7803-0982-0/93, 1993 IEEE, pp. 880-887 (and in U.S. Patent No. 5,434,768). However, the technology, as the author stated, "does not eliminate totally the reverse recovery loss of the diodes." In short, existing converters circuits, while implementing effectively soft switching, suffer from two major defects: the dangerous locked-up mode, which may destroy parts, and the excessive reverse recovery losses of diodes, which compromises power conversion efficiency. EP 0 610 158 A1 discloses a fixed frequency converter including a main switch and an auxiliary switch. A capacitor is electrically coupled in parallel to the main switch, while a diode is electrically coupled in parallel to the auxiliary switch. A reset capacitor is coupled between the main switch and the auxiliary switch. This arrangement allows to transfer energy stored in the fixed frequency converter back to a voltage source connected thereto when the main switch is turned off. As a further result, the main switch can be turned on at zero voltage. Moreover, this fixed frequency converter comprises a secondary tapped winding and a third switching means (saturable reactor, inductor) coupled in series with one section of the tapped secondary winding as well as a first rectifier means coupled in series with the third switching means and a secondary rectifier means coupled between the other section of the tapped secondary winding and the first rectifier means. This arrangement allows to transfer energy through the fixed frequency converter during both on and off time of the main switch. EP 0 471 421 A2 discloses a stabilizing power source apparatus comprising a primary winding, a first and a secondary winding. A first switch is connected in series with one end of the primary winding and a second switch is connected in series with the other end of the primary winding whereby switches are also connected in series to a capacitor being connected in parallel to the primary winding. A first saturable reactor is connected to the first secondary winding and a first rectifying circuit is connected in series with the first saturable reactor. A second saturable reactor is connected to the second winding and a second rectifying circuit is connected in series with the second saturable reactor. The design of the secondary of the stabilizing power source apparatus provides for a decrease of the switching losses. EP 0 474 471 discloses a forward single ended type converter. SUMMARY OF THE INVENTION It is an object of the present invention to provide a DC-to-DC converter circuit which solves these problems associated with the prior art. It yet another object of the present invention to provide a DC-to-DC converter circuit which eliminates reverse recovery losses of rectifier diodes. It is yet an another object of the present invention to provide a DC-to-DC converter circuit which includes a transformer, a main switch and circuitry for automatically transferring the energy stored in the transformer winding back. to the voltage source connected to the primary of the transformer when the main switch is off. It is yet another object of the present invention to provide a DC-to-DC converter system with soft-PWM switching which includes a main switch and an auxiliary switch with lossless snubbers for minimizing the switching losses of the solid state switches. Briefly, the present invention relates to a new forward DC-to-DC converter with soft-PWM switching. The converter presents two pulses to the output low-pass filter within one switching cycle. Hence, power is forwarded to the output twice (double forward) in one switching cycle of the converter. The double forward DC-to-DC converter includes a transformer having a single primary winding and two secondary windings. A main switch, under the control of a PWM control circuit, is connected in series with the primary winding. An auxiliary switch is coupled across with the main switch by way of a resonant capacitor. The resonant capacitor and auxiliary switch are used to automatically transfer energy stored in the transformer winding, while the main switch is off, back to the voltage source connected to the transformer primary. A diode and snubber capacitor is connected across the drain-source terminals for both the main and auxiliary switches for minimizing switching losses. The so-called current mirror is abandoned here by intelligent design of circuit. Hence, the notorious locked-up mode of operation is avoided. The secondary windings of the transformer are connected to a pair of saturable reactors which, in turn, are connected to rectifier diodes. A freewheeling diode is connected across the secondary winding and cooperates with the saturable reactors to limit the load current to the secondary of the converter during the transitions of the main and auxiliary switches. This configuration provides the needed mechanism to eliminate the reverse recovery losses of the rectifier diodes. DESCRIPTION OF THE DRAWINGS These and other objects of the present invention will be readily understood with reference to the following specification and attached drawing wherein: FIG. 1 is schematic diagram of a double forward DC-to-DC converter circuit with soft-PWM switching in accordance with the present invention. FIG. 2A-2M represent the key waveforms of the voltage and current as a function of time of various circuit elements in the DC-to-DC converter circuit illustrated in FIG. 1. FIG. 3A-3E represent the major stages of operation of the double forward DC-to-DC converter circuit in accordance with the present invention, illustrating the direction of current flow during each stage as indicated by arrow signs. FIG. 4 is the schematic diagram of an alternative implementation of the present invention, where the magnetizing current of the XFRM is steered away from the input or voltage source. Hence, the recycled energy is reduced. FIG. 5 is the schematic diagram of yet another alternative implementation of the present invention, particularly suitable for applications where synchronous rectifiers are preferred over conventional rectifier diodes. FIG. 6 is the schematic diagram of the present invention for applications where multiple output voltage are needed. A DETAILED DESCRIPTION OF THE INVENTION Two important considerations in all types of power conversion are efficiency and power density. DC-to-DC converter circuits with soft switching are known to provide relatively better results than conventional PWM switching power supplies. The double forward DC-to-DC converter with soft-PWM switching in accordance with the present invention includes circuitry to minimize switching losses of the switching devices and also eliminates reverse recovery losses associated with the rectifier diodes. An important feature in the double forward DC-to-DC converter circuit in accordance with the present invention is that switching and reverse recovery losses are eliminated with minimal circuit complexity and minimum introduction of losses due to the additional circuitry. The double forward DC-to-DC converter circuit in accordance with the present invention is illustrated in FIG. 1 and generally identified with the reference numeral 20. The forward DC-to-DC converter circuit 20 includes a transformer XFRM having a single primary winding with n 1 turns and dual secondary windings each with equal turns, n 2 . The polarities of the primary and secondary windings are as indicated by the dots in FIG.1. A main switch M 1 , for example, a solid state switch, such as a MOSFET with gate, drain and source terminals, is used to control the energy flow in the primary winding. In particular, the drain and source terminals of the main switch M 1 are connected in series with the primary winding. A pair of input terminals are connected to the primary winding and the main switch M 1 . A source of DC voltage is coupled to the input terminals. An auxiliary switch M 2 , for example, solid state switch, such as a MOSFET with gate, drain and source terminals, is connected with its drain and source terminals electrically coupled across the drain and source terminals of the main switch M 1 . The auxiliary switch M 2 in combination with a resonant capacitor C R is used for resetting the transformer core and for enabling zero-voltage switching for both the main switch M 1 and the auxiliary switch M 2 . The resonant capacitor C R is connected between the two switches M 1 and M 2 as shown. The gate terminals for both the main switch M 1 and auxiliary switch M 2 are connected to control logic such that both switches M 1 and M 2 are operated in a complementary manner wherein both switches are not on at the same time. Also the control circuitry is such that a predetermined dead time is provided after the main switch M 1 is turned off and before the auxiliary switch M 2 is turned on. A dead time is also provided from the time the auxiliary switch M 2 is turned off before the main switch M 1 is turned back on. The dead times are discussed in more detail below. Lossless snubbers C 1 and C 2 are connected in parallel across the drain and source terminals of the main switch M 1 and auxiliary switch M 2 , respectively. These lossless snubbers C 1 and C 2 , as will be discussed in more detail below, reduce the turn off losses of the main switch M 1 and the auxiliary switch M 2 by limiting the voltage across the switches M 2 and M 2 to the voltage V CR across the resonant capacitor C R . The snubbers C 1 and C 2 may be implemented as discrete capacitors, the stray capacitances associated with the switches M 1 and M 2 or a combination of both. Diodes D 1 and D 2 are connected across the drain and source terminals of the switches M 1 and M 2 respectively. As will be discussed in more detail below, the diodes D 1 and D 2 are used, together with the resonant capacitor C R and the magnetizing inductance of transformer XFRM, to minimize turn-on losses of the switches M 1 and M 2 . If the switches M 1 and M 2 are implemented as MOSFETS, the body diodes of MOSFETS may be used for the diodes D 1 and D 2 . Alternatively, discrete diodes for D 1 and D 2 can also be used. A resonant capacitor C R is connected between the respective drain or source terminals of the main M 1 and auxiliary M 2 switches. The resonant capacitor C R in combination with the magnetizing inductance, and the switches M 1 and M 2 automatically transfer the energy stored in the transformer winding back to the DC voltage source connected to the transformer primary winding when the main switch M 1 is turned off as well as enable zero-voltage turn-on of the switches M 1 and M 2 . In particular, once the main switch M 1 is turned off, the energy stored in the transformer core charges the resonant capacitor C R , forward biasing the diode D 2 across the auxiliary switch M 2 to enable the auxiliary switch M 2 to turn on while the diode D 2 is conducting, which, in turn, enables the auxiliary switch M 2 to be turned on at zero voltage. Once the auxiliary switch M 2 is turned on, the energy stored in the transformer winding is automatically returned to a source of DC voltage connected to the primary winding of the transformer XFRM. The resonant capacitor C R also forward biases the diode D 1 to enable turn on of the main switch M 1 while the diode D 1 is conducting thereby allowing the main switch M 1 to be turned on at zero volts. As will be discussed in more detail below, the turn-off voltage stress across the main M 1 and auxiliary M 2 switches is limited to the voltage V CR across the resonant capacitor C R . V CR equals 1.5 V g typically. As mentioned above, the transformer XFRM includes dual secondary windings each having an equal number of turns, n 2 . A pair of rectifier diodes D 3 and D 4 are connected in series with the secondary windings. In order to eliminate reverse recovery losses of the rectifier diodes, D 3 and D 4 , a pair of saturable reactor SR 3 and SR 4 are utilized. In particular, one saturable reactor SR 3 (SR 4 ) is connected in series with each of the rectifier diodes D 3 (D 4 ). A f reewheeling diode D 5 is connected across the input of the low-pass filter. The freewheeling diode D 5 in combination with the saturable reactors SR 3 and SR 4 keeps the load current flowing to the secondary during the transition time of the main switch M 1 and auxiliary switch M 2 . Output terminals are defined across the freewheeling diode D5. A filter consisting of an inductor and a capacitor C may be connected across the output terminals. In order to best understand the operation of the double forward DC-to-DC converter circuit 20 in accordance with the present invention, reference is made to FIG. 3A - 3E which show the four stages of operation of the circuit: forward mode, (FIG. 3A), linear soft transition mode (FIG. 3B), "flyback"/forward mode (FIGs. 3C/3D), and resonant soft transition mode (FIG. 3E) while FIGs. 2A-2M illustrate idealized wave forms of voltages and currents for various components in the circuit. Referring first to FIG. 3A, a forward mode of operation of the converters is illustrated. In this mode, the main switch M 1 is turned on by way of a PWM control circuit 22, while the diode D 1 is conducting, as will be discussed in more detail below. Such PWM controls circuits are known in the art and commonly available. For example, a PWM integrated circuit, model no. SG1843 as manufactured by Silicon General is suitable for such an application. Once the main switch M 1 is turned on, electrical current flows from the positive rail of the DC voltage source through the primary winding of the transformer XFRM, the drain and source terminals of the main switch M 1 and back to the negative rail of the DC voltage source. As will be discussed in more detail below, the main switch M 1 is turned on while the diode D 1 is conducting resulting in essentially a zero-voltage turn-on for the main switch M 1 as shown in FIG. 2I, thus minimizing, if not eliminating, switching losses associated with the turning-on of the main switch of M 1 . After the main switch M 1 is turned on, the electrical current through the primary winding of the transformer XFRM linearly ramps up as a function of the magnetizing inductance of L m of the transformer winding and the input DC voltage V g . During this mode, energy is transferred from the primary winding to the secondary winding and, in turn, to the load connected across the capacitor C. This mode of operation is identical to that of a forward converter. During the forward mode, the saturable reactor SR 3 is saturated and thus turned on allowing current to flow through the diode D 3 , the inductor L, the load and return to the secondary winding of the transformer XFRM. During this condition, the saturable reactor SR 4 is blocking. Immediately after the main switch M 1 is turned off, as illustrated in FIG. 3B, the circuitry enters a soft transition mode. In this mode the saturable reactors SR 3 and SR 4 as well as the freewheeling diode allow for soft switching of the rectifier diode D 3 . In particular, immediately after the main switch M 1 is turned off (during the dead time before the auxiliary switch M 2 is turned on) the voltage V pr across the primary winding transitions from a positive value to a negative value, as illustrated in FIG. 2C. During this period the electrical current I pr through the primary winding of the transformer XFRM is positive as illustrated in FIG. 2C. This electrical current I pr through the primary winding charges the resonant capacitor C R , which, in turn, forward biases the diode D 2 connected across the auxiliary switch M 2 which allows the electrical current I pr to flow through the diode D 2 . During this mode, the snubber capacitor C 1 , connected across the drain and source terminals of the main switch M 1 , is slowly charged to a value V CR , equal to the voltage across the resonant capacitor C R , as shown in FIG. 2I, thus limiting the voltage stress associated with the turn off of the main switch M 1 . While the voltage across the primary winding is greater than zero, the saturable reactor SR 3 conducts thereby biasing the rectifier diode D 3 causing current to flow from the secondary winding through the saturable reactor SR 3 , diode D 3 , inductor L through the load and back to the secondary winding. Once the voltage across the primary winding drops below zero, the freewheeling diode D 5 causes electrical current to circulate through the diode D 5 , inductor L load and back through the diode D 5 . The current is diverted from D 3 to D 5 (the rate of diverting is determined by the secondary leakage). As the current i D3 in the diode D 3 approaches zero, the saturable reactor SR 3 regains its high impedance, eliminating reverse recovery losses for the diode D 3 as indicated in FIG. 2F. Since the saturable reactor SR 4 is still blocking, SR 3 sees no voltage during the transition. Hence, it is soft switched. So is the diode D 3 . Once the auxiliary switch M 2 turns on, the bipolar magnetizing current i mg through the transformer winding swings from a maximum peak to a flat negative peak. While the magnetizing current i mg is positive, the DC converter circuit 20 operates in a "flyback" mode as illustrated in FIG. 3C. Once the magnetizing current i mg goes from a positive value to a negative value while, the auxiliary switch M 2 is closed. When the magnetizing current goes negative, the circuit operates in a forward mode as indicated in FIG. 3D. Referring back to FIG. 3C, in the "flyback" mode, the magnetizing current i mg is circulated from the positive rail of the DC voltage source through the primary winding, resonant capacitor C R , diode D 2 , drain and source terminals of the auxiliary switch M 2 and back to the negative rail of the DC voltage source. The auxillary switch M 2 is turned on while diode D 2 is conducting, realizing zero-voltage switching of the auxiliary switch M 2 . Since the voltage V pr across the primary winding is negative during this period, the saturable reactor SR 3 blocks while the saturable reactor SR 4 conducts. During such a condition the diode D 4 is forward biased allowing electrical current to flow from the secondary winding through the saturable reactor SR 4 , diode D 4 , the inductor L through the load and back to the secondary winding. Note that the energy provided to the load is from the transformer. This is similar to a flyback converter. However, the voltage across the transformer, V pr , is not determined by the output voltage as in a flyback converter. Rather, the voltage V pr is determined by the voltage across the resonant capacitor V CR , which is similar to a forward converter. To indicate this difference, flyback is in quotation marks. Note also that, since the current mirror is abandoned, the duration of this "flyback" mode is much shorter than those in prior art. This is needed to prevent the locked-up mode from onsetting. Once the magnetizing current i mg goes negative, the DC-DC converter circuit 20 operates in a forward mode as illustrated in FIG. 3D causing the magnetizing current i mg to circulate from the negative rail of the DC voltage source through the auxiliary switch M 2 , resonant capacitor C R through the primary winding and back to the positive rail of the DC voltage source. Since the voltage across the primary winding is negative during this condition, the saturable reactor SR 3 blocks while the saturable reactor SR 4 conducts causing current to circulate through the saturable reactor SR 4 , diode D 4 through the inductor L through the load and back to the secondary winding. Note that the energy provided to load is from the resonant capacitor, C R . This energy transfer mechanism is similar to a forward converter. During the "flyback"/forward interval, the voltage presented to the low pass filter is a second voltage pulse, as indicated in FIG. 2M during a switching cycle of the main switch M 1 . The amplitude of the pulse is given by (V g -V CR )/n. This is a significant departure from the conventional forward converter. (In a conventional forward converter, there is only one voltage pulse presented to the low-pass filter). Therefore, the present invention can be called appropriately a double forward converter. Indeed, straightforward analysis yields the voltage conversion ratio is given by which is exactly twice of the value for a forward converter where M = d / n . The feature of double forwarding has demonstrated its ability to improve efficiency anywhere between 2-10%, depending on applications. This gain of efficiency is obtained by realizing a maximum duty ratio close to one, compared to 50% in a conventional forward converter. After the auxiliary switch M 2 is turned off, the DC-DC converter circuit 20 enters second transition mode as illustrated in FIG. 3E. As discussed above, a deadtime exists between the turn-off of the auxiliary switch M 2 and the turn-on of the main switch M 1 . As shown in FIG. 2C, during this deadtime, the voltage on the primary winding V pr ramps up linearly to a positive value equal to the source voltage V g . Immediately after the auxiliary switch M 2 is turned off, electrical current continues to flow from the negative rail of the DC voltage source through the auxiliary switch M 2 , resonant capacitor C R through the primary winding and back to the positive rail of the DC voltage source. Electrical current also flows from the negative rail of the DC voltage source through the snubber capacitor C 1 , thereby discharging the snubber capacitor C 1 to allow the diode D 1 to turn on, which, in turn, enables the main switch M 1 to be turned on at zero voltage during the next cycle. Once the auxiliary switch M 2 is turned off, the snubber C 2 limits the voltage across the auxiliary switch M 2 to a voltage V CR the voltage across the resonant capacitor C R . Note that the negative current in the main switch M 1 needs to be designed to be small. Hence, no significant penalty in added conduction loss is incurred. A small negative current is also consistent with the avoidance of the locked-up mode. After the auxiliary switch M 2 is turned off, the saturable reactor SR 4 conducts, forward biasing the diode D 4 to enable it conduct when the primary voltage V pr is less than zero. Once the primary voltage V pr rises above zero, the freewheeling diode D 5 begins conducting to enable current to circulate through the inductor L through the load and back through the freewheeling diode D 5 while the saturable reactor SR 3 is in an blocking mode thereby providing a soft transition of the rectifier diode D 4 eliminating reverse recovery losses for the diode D 4 . The transition of load current from D 4 to D 5 incurs no switching loss, since the saturable reactor SR 3 is blocking. Note also that during two time durations for soft transitions, the freewheeling diode D 5 limits the load current to the secondary side of the converter. This feature not only allows the primary transition to proceed orderly, but also limits the circulating energy to a minimum (hence, improved efficiency). Another significant improvement of the present invention is its ability to avoid the undesirable locked-up mode. There is no additional circuitry needed. Proper design of the primary parameters allows the circuit to operate in a different mode as indicated in FIG. 2L. The current waveform is no longer symmetrical and the current mirror is abandoned. More time is allowed for the resonant capacitor to reset. Hence, this circuit is tolerable of transient conditions, free of the possibility of being locked up. Consequently, the reliability of the converter is greatly improved. Alternate embodiments of the invention are illustrated in FIGS. 4, 5 and 6. For simplicity, like components are identified with like reference numbers. Primes, double primes etc. are used to distinguish between the various embodiments. All three embodiments operate in a manner similar to the embodiment illustrated in FIG. 1. All three embodiments incorporate feedback from the output terminal of the converter to the PWM control circuit. Other distinctions between each of the embodiments relative to FIG. 1 are discussed below. FIG. 4 is similar to FIG. 1, except for the location of the auxiliary switch M 2 . In this embodiment, the auxiliary switch M 2 is connected across the primary winding n 1 of the transformer XFRM. More specifically, the auxiliary switch M 2 is implemented as a MOSFET having gate, drain and source terminals. The drain and source terminals are electrically coupled across the primary winding. A resonant capacitor C R prime is coupled between the auxiliary switch M 1 and the auxiliary switch M 2 . FIG. 5 represents another alternate embodiment of the present invention. This embodiments is particularly suitable f or applications were synchronous rectifiers are preferred over conventional rectifier diodes. This embodiment is also similar to the embodiment illustrated in FIG. 1 except that the rectifier diodes D 3 and D 4 are substituted with synchronous rectifiers SD 3 and SD 4 . FIG. 6 is another alternate embodiment of the invention which provides for multiple outputs. In this embodiment, the transformer XFRM is provided with a plurality of independent secondary windings. For example, 3 independent secondary windings are illustrated in FIG. 6. As shown, each independent set of secondary windings includes two windings as in FIG. 1. The circuitry otherwise is similar to the embodiment illustrated in FIG.1.

Claims: A DC-to-DC converter circuit (20) comprising: a transformer having a primary winding (n1) and dual secondary windings (n2), said primary winding (n1) adapted to be coupled to a DC voltage source (V g ), a main switch (M1) electrically coupled to said primary winding (n1) and said DC voltage source (V g ), said main switch (M1) adapted to be controlled by a pulse width modulator circuit, a pair of input terminals (+, -) electrically coupled to said primary winding (n1) and said main switch (M1), a first capacitor (C1) connected in parallel across said main switch (M1), a reset circuit electrically coupled to said primary winding (n1) and said DC voltage source (V g ) for automatically returning magnetizing current to said DC voltage source (V g ) when said main switch (M1) is turned off, said reset circuit including an auxiliary switch (M2) electrically coupled to said primary winding (n1) and the terminal (-) of said DC voltage source (V g ) and a second diode (D2) connected in parallel across said auxiliary switch (M2), a resonant capacitor (C R ) electrically connected between said main and auxiliary switches (M1, M2), a first saturable reactor (S r3 ) electrically coupled to one of said dual secondary windings (n2) and serially coupled to a first rectifier device (D3), a second saturable reactor (S r4 ) electrically coupled to the other of said dual secondary windings (n2) and serially coupled to a second rectifier device (D4), and a freewheeling diode (D5) electrically coupled between the mid point of said dual secondary windings (n2) and one of said first and second rectifier devices (D3, D4), characterized by a first diode (D1) connected in parallel across said main switch (M1), and a second capacitor (C2) connected in parallel across said auxiliary switch (M2). A DC-to-DC converter circuit as recited in claim 1, including: at least one further dual secondary windings (n21, n22, n23), a first saturable reactor (S r31 , S r32 , Sr 33 ) respectively electrically coupled to one of said at least one further dual secondary windings (n21, n22, n23) and serially coupled to a respective first rectifier device (D31, D32, D33), and a second saturable reactor (S r41 , S r42 , S r43 ) respectively electrically coupled to the other of said at least one further dual secondary windings (n21, n22, n23) and serially coupled to a respective second rectifier device (D41, D42, D43), and a respective freewheeling diode (D51, D52, D53) electrically coupled between the mid point of said at least one dual secondary windings (n21, n22, n23) and one of said respective first and second rectifier devices (D31, D32, D33, D41, D42, D43). A DC-to-DC converter circuit as recited in claim 1 or 2, wherein said main switch (M1) and/or said auxiliary switch (M2) are solid state switches. A DC-to-DC converter circuit as recited in claim 3, wherein said solid state switch (M1, M2) is a MOSFET having gate, drain, and source terminals. A DC-to-DC converter circuit as recited in one of the claims 1-4, wherein said first capacitor (C1) and/or said second capacitor (C2) are snubber capacitors. A DC-to-DC converter circuit as recited in one of the claims 1-5, wherein said dual secondary windings (n2) have equal turns. A DC-to-DC converter circuit as recited in one of the claims 1-6, wherein said first and second rectifier devices (D3, D4) are synchronous rectifier. A DC-to-DC converter circuit as recited in one of the claims 1-6, wherein said first and second rectifier devices (D3, D4) are rectifier diodes. A DC-to-DC converter circuit as recited in one of the claims 1-8, wherein a pair of output terminals is defined across said freewheeling diode (D5), and a filter circuit is electrically coupled to said pair of output terminals. A DC-to-DC converter circuit as recited in claim 9, wherein said filter circuit includes an inductor (L) and a capacitor (C).

Inventors: TAN F DONG, TAN, F. DONG
Applicants: NORTHROP GRUMMAN CORP, NORTHROP GRUMMAN CORPORATION
Application year: 1997
Main classification: H02M3
IPC codes: H02M3