IMPROVED METHOD AND APPARATUS FOR CONTROLLING A SWITCHED RELUCTANCE MACHINE

Abstract: A method and apparatus for controlling the average voltage applied to a phase winding of a reluctance machine during a period of current increase, controlling the current when a desired current is reached, and controlling the voltage applied during a period of current decrease, to improve machine efficiency and thereby reduce acoustical noise produced by the machine, and unwanted vibration. The apparatus comprises circuitry and associated elements which are operable to switch the phase winding into circuits which apply either a positive DC voltage, a zero voltage, or a negative DC voltage, to increase, maintain, or decrease the current in the winding. The timing and the rate of the switching is controlled as a function of the angular position and speed of a rotor of the reluctance machine, the torque output of the machine, and the desired speed or torque. A method is provided whereby a duty cycle corresponding to a specific reluctance machine, as well as to specific operating parameters of the machine at a given rotor position, speed, and torque, is calculated by the controller for applying voltage to the phase winding during the period of current increase to increase the current from at or near zero to a desired peak current. Additionally, a method is provided for calculating a duty cycle for the time at which the phase is de-energized, corresponding to a specific reluctance machine, and to specific operating parameters of the machine at a given rotor position, speed, and torque, as well as a method for changing the duty cycle to drive the current in the winding to zero or near zero. The calculation of the appropriate duty cycle for the specific operating conditions of the reluctance machine controls the net average voltage applied, has been found to reduce abrupt changes in the current. This reduction correlates into reduced abrupt changes in the magnetic flux, resulting in reduced acoustic noise and unwanted vibration.

Description: IMPROVED METHOD AND APPARATUS FOR CONTROLLING A SWITCHED RELUCTANCE MACHINE BACKGROUNDOF THE INVENTION In general a reluctance machine is an electric machine in which torque is produced bv the tendency of its movable pan to move into a position where the reluctance of an excited winding is minimized (i e the inductance is maximized) In one tvpe of reluctance machine the energization of the phase windings occurs at a controlled frequency These machines are generally refened to as synchronous reluctance machines In a second type of reluctance machine circuitry is provided for detecting the angular position of the rotor and energizing the phase windings as a function of the rotor's position This second type of reluctance machine is generally known as a switched reluctance machine Although the description of the present invention is in the context of a switched reluctance machine, the present invention is applicable to all forms of reluctance machines, including synchronous and switched reluctance motors, synchronous and switched reluctance generators, as well as to other machines that have phase winding arrangements similar to those of switched reluctance machines The general theory of design and operation of switched reluctance machines is well known and discussed, for example in The Characteristics. Design and Applications of Switched Reluctance Motors and Drives, by Stephenson and Blake and presented at the PCIM '93 Conference and Exhibition at Nuremberg, Germany, June 21-24, 1993 When a switched reluctance machine is running, including at low speeds or a standstill, the torque (and other machine perfoπnance parameters) may be adjusted by monitoring the rotor's position, energizing one or more phase windings when the rotor is at a first angular position, referred to as the "turn-on angle (T 0N ,)" and then de-energizmg the energized windings when the rotor rotates to a second angular position, referred to as the "tum-off angle (T OFF ) " The angular distance between the turn-on angle and the tum-off angle is often referred to as the "conduction angle " At standstill and at low speeds, the torque of a switched reluctance machine can be controlled by varying the magnitude of the cunent in the energized phase windings over the period defined by T 0N and T 0FF Such current control can be achieved by chopping the current using a current reference with phase current feedback Such current control is referred to as "chopping mode" current control Alternately, pulse width modulation (PWM) voltage control may be used As the angular speed of the motor increases, a point is reached where the amount of current which can be delivered into a phase winding during each phase period is limited by the rapidly increasing inductance and counter emf associated with the winding At such speeds pulse width modulation or chopping strategies are less desirable and the torque of the machine is commonly controlled by controlling the duration of the voltage pulse applied to the winding during the phase period with respect to the rotor's position Because a single pulse of voltage is applied during each phase period this form of control is often referred to as "single-pulse control " As a switched reluctance motor (or generator) operates, magnetic flux is continuously increasing and decreasing in different pans of the machine This changing flux will occur in both chopping mode and smgle-pulse current control The changing flux results in fluctuating magnetic forces being applied to the ferromagnetic parts of the machine These forces can produce unwanted noise and vibration One major mechanism by which these forces can create noise is the ovahzing of the stator caused by magnetic forces normal to the air-gap Generally as the magnetic flux increases along a given diameter of the stator the stator is pulled into an oval shape by the magnetic forces As the magnetic flux decreases, the stator pulls or springs back to its undistoned shape. This ovalizing and springing back of the statorwill produce audible noise and can cause unwanted vibration. In addition to the distortions of the stator by the ovalizing magnetic forces, acoustic noise and unwanted vibration may also be produced by abrupt changes in the magnetic forces in the motor. These abrupt changes in the gradient of the magnetic flux (i.e.. the rate of change of the flux with time) are referred to as " hammer blows'' because the effect on the stator is similar to that of a hammer strike. Just as a hammer strike may cause the stator to vibrate at one or more natural frequencies (determined by the mass and elasticity of the stator) the abrupt application or removal of magnetic force can cause the stator to vibrate at one or more of its natural frequencies. In general, the lowest (or fundamental) natural frequency dominates the vibration, although higher harmonics may be emphasized by repeated excitation at the appropriate frequency. In addition to the stator distortions resulting from the ovalizing and hammer blow phenomena described above, the fluctuating magnetic forces in the motor can distort the stator in other ways, as well as distorting the rotor and other parts of the machine system. For example, distortions of the rotor can cause resonance of the rotor end- shields. These additional distortionsare another potential source of unwanted vibration and noise. Although the problem of unwanted acoustic noise and vibration has been recognized, known control systems for reluctance motors do not adequately solve the problem. For example, the general problem of acoustic noise in switched reluctance motor systems is discussed in CY. Wu and C. Pollock, "Analysis and Reduction of Vibration and Acoustic Noise in the Switched Reluctance Drive," Proceedings of the IAS '93 pp. 106-113 (1993). In general, the method suggested by Wu and Pollock involves control of the current in the phase winding such that the current is controlled in two successive switching steps with the second switching step occurring approximately one-half of a resonant cycle after the first where the resonant cycle is defined by the natural frequency of the machine. This approach is typically implemented by switching off one of the power devices at a first point in time to cause a fust stepped reduction in applied voltage, and then later switching off the second power device. Between the time when the first switching device is switched off and the second switching device is switched off, the current is allowed to freewheel through a freewheelingdiode and the second switching device. The two-step voltage-reduction approach to noise reduction in switched reluctance motors discussed above suffers from several limitations and disadvantages. One such limitation is that in many cases the two-step voltage- reduction approach requires precise switching of the switching devices within the interval defined by the tum-on and turn-off angles (i.e., the angular interval during which the phase winding is energized). Still further, the two-step voltage-reductionapproach limits the flexibility to dynamically adjust the freewheeling period for each phase cycle. As discussed above, in the two-step voltage-reduction approach, the duration of the freewheeling period is optimized to reduce the noise produced by the system at a single fundamental frequency. There are many instances when it would be desirable to optimize the freewheeling duration according to other criteria. An additional limitation of the two-step voltage-reduction approach, and other approaches that utilize freewheeling to reduce noise, is that, since there is typically only one freewheeling period per phase energization cycle, freewheeling generally reduces noise produced by only a single frequency of the motor system. Freewheeling to reduce noise at one frequency does not reduce noise produced at other frequencies, in motor systems that have multiple resonant frequencies. Accordingly, such approaches do not reduce many sources of unwanted noise. A further disadvantage with the freewheeling approaches is that there are several motor control systems (e g , H-circuits with a split capacitor, third-rail circuits, ring circuits and the like) that simply do not allow freewheeling These systems cannot use freewheelmgto reduce noise The present invention overcomes many of the limitations and disadvantages associated with known systems and provides a unique method and apparatus for controlhngthe phase voltage and the phase winding currents in a phase windings of a switched reluctance machine to, for example, reduce unwanted machine noise and vibrations SUMMARY OF THE INVENTION One aspect of the present mvention concerns a method and apparatus for controlling the average voltage applied to a phase winding of a machine, particularly a reluctance machine, during a period of current increase, and a period of current decrease, to improve machine performance by reducing abrupt changes in the phase current and the magnetic flux associated with the phase or with one or more adjacent energized phases One aspect of this improved performance is a reduction in audible noise and vibration generated by the machine Audible noise and vibration are generated in a reluctance machine in part as a result of rapid changes in the magnetic flux which tend to distort or ovaiize machine parts Controlling the rate of change in the voltage and current in phase windings duπng the penods when a winding is energized from a zero or near-zero voltage state to a maximum voltage state, or from a maximum voltage state to a zero or near-zero voltage state, can result in less rapid change in the magnetic flux, thereby reducing audible noise and vibration An exemplary apparatus of the present mvention concerns a controller and associated circuitry The method of the present invention concerns a series of steps performed by the controller which govern the duty cycle of voltage pulses applied to a phase winding durmg current increase and current decrease, as a function of the speed and torque output and requirements of the reluctance machine The controller contains circuitry which can operate in several distinct modes The different modes may be selected based on the output requirementsofthe system, e g speed or torque requirements The controller may operate in a smgle mode, or may be programmed to change modes based on the actual operating characteristics of the machine The different modes define the optimum rotor angular positions at which the phase windings are energized and de- energized, in accordance with the desired speed and/or torque performance of the machine The controller also contains circuitry which controls the voltage applied to the phase at the time the windings are energized, until a desired maximum current is reached The circuitry may mclude a microprocessor which determines the maximum current desired in the phase winding based on feedback signals representing the actual speed and torque of the machine In one exemplary embodiment, from the time when the phase is energized, until the time when the maximum current is reached in the phase, the average voltage applied to the windmg is controlled using soft-chopping techniques, I e , switching the phase windmg between a circuit which applies a positive DC voltage and a circuit which applies a zero voltage, thereby allowing the current in the phase winding to freewheel and decrease slowly In this embodiment, the soft-chopping duty cycle is calculated as a function of the angular speed of the rotor and the torque generated by the machine, and the desired speed or torque of the machine As a result, the gradient of the increasing current in the phase winding will vary with the speed and torque of the machine In another embodiment, from the time when the phase is at its desired maximum cuπent, until the time when the phase is de-energized, soft-chopping techniques are continued to maintain the cunent at the desired level Alternative methods of controlling the current level can be applied, such as hard-chopping techniques, whereby the phase is switched between a circuit which applies a positive DC voltage and a circuit which applies a negative DC voltage, thereby quickly driving the current in the winding to zero In an exemplary embodiment, at the tune when the phase is de-energized, until the time when the current in the phase is at or near zero, the average voltage applied to the winding is controlled by using hard-chopping techniques An aspect of this embodiment is to calculate an initial duty cycle for the applied voltage pulses which is a function of the last soft-chopping duty cycle, and to apply a net average voltage to the phase winding in the first hard-chopping duty cycle which is the same as the net average voltage at the time the phase was de-energized It is also an aspect of this embodiment that the duty cycle of the hard-chopping pulses, duπng the time period when the current in the phase is decreasmg, is modified and controlled such that the net average voltage applied to the winding changes from a positive DC voltage, to zero voltage, to a negative voltage, until the current in the windings reaches zero or a desired value near zero The rate of change of the duty cycle varies as a function of the angular speed of the rotor and the torque generated by the machine, and the desired speed or torque of the machine As a result, the gradient of the decreasmg current tn the phase winding will vary with the speed and torque of the machine In other exemplary embodiments, where the current in the phase prior to de-energization is controlled by either hard chopping or soft chopping, the last complete duty cycle or net average voltage prior to de-energization is captured and the same hard chopping or soft chopping switchmg strategy is continued with the duty cycle ramped down from a positive net average voltage to zero or a net negative voltage Thereafter, at one or more predetermined times or at a one or more preselected rotor positions durmg the period where the duty cycle is ramped down, the duty cycle may be captured agam, and the switchmg strategy may be changed again This could mclude changmg switchmg strategies such that a hard chopping duty cycle is applied, starting with the same net average voltage as the prior duty cycle but ramped down at a different rate, or actuating both switches into a non-conducting state thereby applying the full negative voltage It will be evident to one skilled m the art that the controller of the present mvention can combine the various possible switching strategies during the period after the phase winding is de-energized in many possible combinations other than those discussed above The controller of the mvention can perform these techniques for controlling the rate of current increase and current decrease in a single phase winding of a reluctance machine or other device m which current is controlled as a function of a rotor position, or in multiple phase windmgs The control apparatus and methods are applicable to any method of operation, and any method of determining the rotor positions at which the phase is energized and de- energized, smce the rotor speed and machine torque are factored into the duty cycle of the applied voltage pulses required BRIEF DESCRIPTION OF THE DRAWINGS Other aspects and advantages of the present invention will become apparent upon readmg the following detailed description and upon reference to the drawings in which Figure 1 illustratesa reluctance machine system in accordance with the present invention Figure 2 illustrates in greater detail the reluctance machine 20 of Figure 1 Figure 3A illustrates in greater detail the power converter30 of Figure I Figure 3B illustrates the types of voltages that may be established across a phase winding in a reluctance machine by power connector 30 Figures 4A and 4B illustrate in greater detail the controller 40 of Figure 1 Figure 5A illustrates in flowchart form a method used by controller 40 for selecting a hard-choppmg duty cycle for chopping at the T 0FF rotor position Figure 5B illustrates a typical maximum applied average voltage (defined by a simplified HARD-RAMP- START duty cycle) vs torque output calculation for vaπous speeds Figure 5C illustrates voltage applied to a phase windmg by the method and apparatus of the current invention and the resulting current waveform generated in the phase Figure 5D illustrates the net average voltage applied to a phase winding by the method and apparatus of the current mvention and the resulting current waveform generated in the phase Figure 6A illustrates in flowchart form one method in which controller 40 performs a controlled RAMP- DOWN of the phase current by gradually reducing the net average voltage applied to the phase wing Figure 6B illustrates exemplary curves for selecting a RAMP-DOWN-GRADIENT, which defines the rate of change in the net average voltage applied to the winding, as a function of machine speed and torque Figure 6C illustrates exemplary circuitry for implementing the RAMP-DOWN method of the present invention Figure 7A illustrates an alternate embodiment of the invention wherem a +HVDC to zero voltage soft- chopping switching strategy is employed at T OFF , the duty cycle is ramped down to zero, and then a hard-choppmg switchmg strategy is employed Figure 7B illustrates another alternate embodiment of the mvention wherem a +HVDC to zero voltage soft- chopping switchmg strategy is employed at T OFF , the duty cycle is ramped down to zero, and then a -HVDC to zero voltage soft-choppingswitchtng strategy is employed Figure 8A illustrates in flowchart form a method in which controller 40 performs ( 1 ) a controlled RAMP-UP of the phase current and (n) efficient fixed frequency chopping circuit in the active cycle Figure 8B illustrates current in a phase windmg and the duty cycle of the applied voltage pulses duπng RAMP-UP Figure 8C illustrates exemplary circuitry for implementing the RAMP-UP and RAMP-DOWN control methods of the present invention Figures 9 A and 9B compare a traditional current waveform (Figure 9A) with a current waveform generated through use of the methods and apparatus of the present invention (Figure 9B) Similar reference characters indicate similar parts throughout the several views of the drawings DETAILED DESCRIPTION OF THE INVENTION Although the present invention is discussed in terms of a system which includes a reluctance machine, the method and apparatus may be applied to many other machine systems in which current is applied to a winding or coil as a function of a rotor or armature position, including various inductance motor systems, stepper motor systems, and other motor or generator systems Turning to the drawings and referring to Figure 1. one example of a system 10 that may be used to implementthe methods of the present invention is illustrated in block form As illustrated, an electronic controller 40 receives signals from a rotor position transducer ("RFT*) 3 The RPT 3 detects the angular position of the rotor of a reluctance machine 20 and provides a controller 40 signals representative of the rotor's position RPT 3 may comprise optical or magnetic sensor(s) and may be of conventional construction Embodiments are envisioned wherein the RPT is replaced by circuitry that monitors the electrical characteristics of the phase winding and provides signals representative of the rotor's angular position and/or speed without the use of position sensors One such sensorless approach is disclosed in the currently pending application Serial No 08/549457, "Rotor Position Sensing in a Dynamoelectπc Machine Using Coupling Between Machine Coils' filed November 27, 1995, assigned to the assignee of the present invention In the embodiment of Figure 1 the controller 40 derives an indication of the angular speed of the rotor of the reluctance machine 20 by appropriately processing the position information provided by the RPT 3 Alternate embodiments are envisioned wherem a separate tachometer or similar device provides speed information to the controller 40 In addition to receiving signals from RPT 3 conceming the rotor's position and angular speed of the machine, controller 40 also receives via line 4 feedback signals from a power converter 30 In one embodiment, the feedback signals represent the currents that are supplied by the power converter 30 to the phase windings of the rotor The controller also receives an externally generated signal on line 2 correspondmg to the required torque or speed of the machme 20 In response to the rotor position signals from RPT 3, the feedback signals from a power converter 30 and the torque/speed command on line 2, the controller 40 provides switchmg signals (sometimes referred to as "firing signals") via lmes 4a-4c to a number of power switching devices that comprise a power converter 30 The switchmg devices in the power converter 30 are connected via lmes 5a-5c to three phase windings A, B and C of a three phase switched reluctance machine 20 The three phase windings of the machme (A, B, and C) are schematically illustrated in Figure 1 Those skilled m the art will recognize that although a three phase machme has been shown for illustrative purposes, machines with more or less than three phase wmdmgs can be used The present invention applies equally to such machines In general, the electronic controller 40 responds to the position signals from the RPT 3 by generating firing signals for each of the three phases of the motor to switch the power devices in power converter 30 such that the phase windings A, B and C are energized in the proper sequence over angular periods of rotor rotation to produce torque in a desired direction Figure 2 illustrates in greater detail the three phase reluctance machine 20 of Figure 1 In general the reluctance machme 20 consists of a stator 22 that is constructed accordmg to known techniques from a stack of stator laminationsthat define twelve inwardly projectmg stator poles 23 These poles define a principal stator axis (extending outwardly from Figure 2) A rotor 24 is coupled to a rotatable shaft (not shown) that is co-axial with the principal axis of the stator The shaft is mounted on bearings and is free to rotate The rotor 24 is formed from a stack of rotor laminationsthat define eight outwardly projecting rotor poles 25 Associated with each stator pole 23 is a wound coil of wire 26a, 26b and 26c The coils of opposing stator poles are placed such that current flowing in the opposing stator poles at the same time will establish inwardly pointing electro-magneticsof opposite polarities In the reluctance machine of Figure 2, sets of coils associated with opposing stator poles 23 are coupled together to form three phase windings A, B and C where each phase winding is comprised of four coils 26 and where each phase is associated with four stator poles 23 The terminating ends of the three phase windings A B and C are coupled to the output of power converter 30 via connection lines 5a, 5b and 5c When electric current flows though the phase windings of a given phase (e g , phase winding A) two sets of opposing elecrro-magneticsof opposite polarity will be established in the machine This is illustrated in Fιgure 2 by the references to Phase A N0RTH and Phase A S0UTH electro-magnets which will be established when electric current is flowing in a given direction in the phase winding A The electro-magnets established by this current will produce a force of attraction between the energized stator poles 23 and the rotor poles 25 which will produce a torque By switching energization from one phase winding to another, the desired torque may be mamtained regardless of the angular position of the rotor By switching the energization of the phase windings to develop positive torque, the machine may be operated as a motor, by energization of the phase windings to develop a negative torque the machine may be operated as a brake or generator The magnitude of the produced torque can be controlled by controlling the current in the energized phase winding which controls the strength of the establishedelectro-magneticfield For the sake of illustration, a three phase machme having twelve stator poles and eight rotor poles (i e a 12/8 machine) has been shown Those skilled in the art will recognize that other combinations (e g , 6/2, 6/4, etc ) are possible and that machines with more or less than three phase wmdmgs can be used The present mvention applies equally to such machines Moreover, the present mvention is applicable to inverted machines, where the stator is positioned withm the bore of an outer rotating rotor, and to linear machines, where the rotor moves linearly with respect to the stator As discussed in connection with Figure 1 , the phase windings of the reluctance machine are energized by the application of power to the phase windings by power converter 30 Power converter 30 is illustrated m greater detail in Figure 3 A Turing to Figure 3 A, AC power is supplied to the power converter 30 by input power lmes AC+ and AC- An earth ground may also be applied to the power converter 30 via a GND line An appropπately sized electro-magnetic interference ("EMI") filter 31 receives and conditions the mput AC power before supplymg it to a full-wave rectifier 32 Although not illustrated in Figure 3A, front-end power protection circuitry may be used to prevent damage to the components of the power converter 30 For example, a fuse rated to the maximum allowable amperage may be placed m series with one of the AC+ or AC-power lmes or a motor thermal cut off may be coupled to the EMI filter 31 to cut power off to the reluctance machine 20 in the event that excessively high temperatures are detected in the vicinity of the machme 20 or the power converter 30 The full-wave rectifier 32 receives the AC power and converts in to DC power such that high voltage DC power is available on high voltage DC bus lines +HVDC and HVDC Commoπ Parallel connected DC bus capacitors 34 are used to filter the output of the full wave rectifier 32 and to provide a DC voltage across the high voltage DC bus While the absolute magnitude of the DC voltage provided across the DC bus will vary depending of the power rating of a given power converter the DC voltage across the high voltage DC bus can reach levels of 160 Volts for 1 10 Volt AC input and over 300 Volts for 220 Volt AC input Auxiliary DC power supplies consisting of a string of appropriately sized resistors may be driven from The high voltage DC bus In the example of Figure 3 A a 5 Volt Vcc signal for powering the controller 40 is provided by resistor chain 33a A similar resistor chain 33b provides a source of 5 Volt power to the RPT 3 Although not illustrated in Figure 3 A additional EMI or appropriate under voltage detectors (e g , a resistive chain with a resistive/capacitive load) could be coupled across the high voltage DC bus Such elements are not particularly related to the subject matter of this disclosure and are within the understanding of one of ordinary skill in the art and are thus not discussed in detail herein Coupled across the high voltage DC bus are power switching devices and diode "groups'' where each group is associated with one of the three phases of the reluctance machine 20 Specifically, each group includes an upper power switchmg device 35, a lower power switchmg device 36, an upper fly-back diode 37 and a lower fly¬ back diode 38 A current measuring sense resistor arrangement 39 is associated with each of the power switchmg device groups These current measurmg sense resistor arrangements 39a, 39b and 39c provide voltages that corresponds to the current flowing through the phase wmdmgs associated with each arrangement As discussed more fully below, these "current feedback" signals may be used by the controller 40 to control the current in the phase windings In the embodiment of Figure 3A, each of the current measuring sense resistor arrangements 39a, 39b and 39c consists of a number of parallel connected resistors. This arrangement is beneficial in some applications because it allows for the use of lower power rated (and thus less expensive) resistors A smgle resistor of appropriate value could also be used. The power switchmg devices 35 and 36 may comprise any suitable power switching devices such as MOSFETs, IGBTs, bi-polar transistors, SCRs or a combmation of the above For example, if only the upper switches will be modulated for current control, then fast switchmg power MOSFETs can be used for the upper switching devices 35 and slower switchmg (but lower loss) IGBT devices can be used for the lower power switchmg devices 36 By controlling the actuation of the power switchmg devices 35 or 36, different voltage potentials can be established across the phase winding to which the appropriate group is coupled Figure 3B illustrates the types of voltages that may be established across a phase windmg coupled to one of the power switching groups of Figure 3A Using the Phase A group as an example, Figure 3B(I) illustrates the voltage that will be established across Phase wmdmg A when both the upper power switchmg device 35a and the lower power switching device 36a are actuated into a conducting (or "closed") condition When switches 35a and 36a are so actuated, the phase winding is coupled across the high voltage DC bus and (ignoring losses introduced by the switching devices) the voltage across Phase A winding is substantially equal to the high voltage DC bus voltage (+HVDC) Figure 3B(1I) illustrates the voltage that will appear across the Phase A winding when both of the power switching devices 35a and 36a are actuated into a non-conducting (or "open") configuration when current is flowing in Phase winding A Because of the large inductance of the Phase winding, the current in the winding cannot instantaneously change and the current will thus continue to flow through the winding via the fly-back diodes 37a and 38a Because both the upper and lower flv-back diodes are conducting in this arrangement, the voltage applied across Phase winding A winding is substantially the negative of the high voltage DC bus voltage (-HVDC) Figure 3B(III) illustrates the voltage that will appear across Phase winding A when either the upper or the lower power switching devices 35a or 36a (but not both) is actuated into a non-conducting state while the other is in a conducting state, and cunent is flowing in the phase wmdmg In this arrangement, because the cunent cannot instantaneouslv change the cunent will continue to flow through the conducting power switching device and through the flv-back diode associated with that power conducting device For example, if the lower power switchmg device 36a is actuated to a non-conductive state, current will continue to flow through the upper power switching device 35a and through the upper fly-back diode 37a Similarly if the upper switching device 35a is actuated to a non-conductive state, current will continue to flow through the lower power switchmg device 36a and lower fly-back diode 38a In either arrangement, the total voltage across the Phase A winding (agam ignoring losses introduced by the switchmg devices and diodes) will be approximately zero volts This arrangement, where current is circulating through the windmg with approximately zero volts across the winding, is sometimes referred to as a "freewheeling" arrangement smce the current "freewheels" through the wmdmg By controlling the actuation of the power switchmg devices it is possible to implement various switchmg schemes For example, as illustrated in Figure 3B(IV), the upper and lower switchmg devices can both be alternately actuated to be conductive and non-conductive, simultaneously causmg the voltage across the phase wmdmg to vary from +HVDC to -HVDC This scheme is sometimes referred to as "hard-choppmg " By appropriately selecting the ratio of time for which the +HVDC voltage is applied to the phase windmg as opposed to the -HVDC voltage, the net average voltage applied to the phase wmdmg may be either positive or negative such that the current m the phase wmdmg may be controlled to mcrease or decrease Figure 3B(V) illustrates an alternate switchmg scheme that may be used to control the average voltage applied to the phase windmg In this switchmg scheme, either (l) the upper power switchmg device 35a is left in a conducting state while the lower power switchmg device 36a is alternately switched from a conductmg to a non- conductmg state or (n) the lower power switchmg device is left in a conductmg state while the upper switchmg device 35a is alternately actuated between conductive and non-conductive states In addition, the power switchmg device that is left in a conducting state and the power switchmg device that is switched between a conductmg and a non-conducting state, can be alternated For example, the upper power switching device 35a, may be left in the conducting state for one cycle while the lower power switchmg device 36a is alternately opened and closed, and the following cycle the lower power switchmg device 35a, may be left in the conductmg state while the upper power switchmg device 36a is alternately opened and closed Alternately, freewheeling may be implemented on other than a per cycle basis This altematmg of the power switchmg devices results in reduced switchmg frequency, extended service life of the devices, reduced switching and system losses, and allows the use of slower switchmg devices In this switching scheme the voltage applied across the phase winding varies from +HVDC to 0 Volts This type of switching scheme, where the voltage varies from a given value to zero, is sometimes referred to as "soft-chopping ' In the soft-chopping scheme of Figure 3B(V) where the voltage across the phase winding vanes from +HVDC to 0 the net average voltage applied to the phase winding can be controlled to be a desired positive but not a negative value Figure 3B(VI) illustrates a soft-chopping scheme in which, while cunent is flowing in the phase winding, either: (i) the upper power switching device 35a is left in a non-conducting state while the lower power switching device 36a is alternately switched from a conducting to a non-conducting state or (ii) the lower power switching device is left in a non-conducting state while the upper switching device 35a is alternately actuated between conducting and non-conducting states. The alternating of the power switching device that is left in a non¬ conducting state and the power switching device that is switched between a conducting and a non-conducting state, discussed above, applies equally in this switching scheme with the same resulting benefits. In this switching scheme the voltage applied across the phase winding varies from -HVDC to 0 Volts. In the soft-chopping scheme of Figure 3B(VI) where the voltage across the phase winding varies from -HVDC to 0 the net average voltage applied to the phase winding can be controlled to be a desired negative, but not a positive, value. The particular switching scheme used for actuating the power switching devices 35a-c and 36a-c in the power converter of Figure 3A is determined by the switching signals applied to the gates of the power switching devices by the controller 40. Controller 40 is illustrated in greater detail in Figures 4A and 4B. Referring to Figure 4A, the controller 40 comprises a semiconductor integrated circuit chip 41 (indicated by the dashed box) and associated external circuitry. In the embodiment of Figure 4A the semiconductor integrated circuit chip 41 comprises a digital core 42 that is surrounded by additional circuit elements including comparators 43 and 44a-44c. Comparators 43 and 44a-44c are "digital comparators" in that the output of the comparators is either a logic level one (when the voltage at the + input to the comparator is greater than the voltage at the - input to the comparator) or logic level 0 when the converse is true. Control chip 41 may be an integrated circuit chip that includes a properly programmed microprocessor or microcontroller. In one embodiment, the control chip 41 consists of an application specific integrated circuit chip ("ASIC") that is coupled to an external electrically erasable programmable read only memory ("EEPROM") that includes operational data and instructions from which a controller within the ASIC can operate. While the embodiment of Figure 4 illustrates many of the components of the controller as located on chip 41 the components of the controller could be placed on a number of discrete chips, or analog circuits could be used. The control chip 41 receives as input the output of the RPT 3 that indicates the angular position of the rotor. The precise form that the RPT input takes will vary depending on the specific type of RPT used to detect the rotor position. For example, the RPT input could consist of a digital word representing the actual position of the rotor (if a digital position encoder is used as an RPT) or simply pulses that indicate that the rotor has rotated through a preestablished angular period. In any event, the various possible forms of the RPT and the reception of RPT signals to provide both rotor position and angular speed information will be well known to one of ordinary skill in the art, and will not be addressed herein in detail. In addition to receiving the RPT signals as an input, the control chip 41 also receives via line 2 the externally derived signal representing the desired speed or torque of the machine. For purposes of discussion it will be assumed that the externally applied signal on line 2 is a representation of the desired torque output of the machine although those skilled in the art will appreciate that the control methods and apparatus discussed herein are also applicable to speed control signals. The externally generated signal representing the desired torque output of the machine is received by the chip 41 via line 2 and, in Figure 4A, is compared to a reference voltage VREF in comparator 43 to produce a modified torque demand signal on line 2 In the embodiment of Figure 4A it is assumed that the externally supplied torque demand signal is a pulse width modulated (PWM) signal where the duty cycle of the PWM signal conesponds to the ratio of the demanded torque to the maximum torque available from the machine In this embodiment the magnitude of the V R£F voltage signal is set to a level that is between the logic zero and logic one levels that define the PWM signal Thus, the signal appearing on line 2' is a PWM torque demand signal that has a duty cycle that corresponds to the duty cycle of the externally supplied PWM torque demand signal received on line 2 For certain systems, the externally supplied torque demand signal on line 2 will not be a PWM signal but will instead be an analog signal whose voltage magnitude corresponds to the desired torque output of the system The controller of the present mvention is capable of receivmg and operatmg from such signals with little or no modification A circuit that would enable the controller 40 of the present mvention to receive an analog signal is illustrated in greater detail in Figure 4B In Figure 4B only the portion of Figure 4A relevant to the reception of the externally supplied torque demand signal is illustrated In general, the circuitry of Figure 4B is the same as that of Figure 4A except that a resister/capacitor ("RC") filtering network 45 is coupled to the inverter output of comparator 43 For purposes of illustration the inverted output of comparator 43 is illustrated as being supplied by an inverter 46 This inverted signal may be alternately generated in the digital core 42 of control chip 41 and provided via an external pm When an analog signal is received on line 2 that is greater than the voltage at the non-inverted terminal of comparator 43, the inverted output of the comparator 43 will be logic high or positive This positive voltage will cause elecrπcal current to flow mto the RC network 45, charging the capacitor at a rate defined by the RC time constant of network 45 Eventually, the voltage across the capacitor (which is applied to the non-inverting terminal) will exceed the input analog voltage of control chip 41 and the output of converter 43 will switch states, causing the inverted output to go to logic zero (ground) thus draming the charge off of the capacitor Eventually the voltage across the capacitor will fall below that of the analog voltage applied on line 2 and the output of the converter 43 will agam switch states In one embodiment, the signal on line 2' is applied as a control signal to an up/down counter that is clocked at a given frequency, such that the counter counts up when the signal on line 2 is logic high and counts down when the signal is logic low In this embodiment, the output at the counter corresponds to the speed or torque command signal Refemng again to Figure 4 A, the digital core 42 of the controller 40 receives the RPT information and the PWM torque command on line 2 and, based on this information, determine the appropriate firing signals to be supplied via lines 5a, 5b and 5c to the power switching devices of power converter 30 The precise manner in which the control circuit generates the firing signals depends on the control mode in which the control chip is operating The control chip 41 is capable of operating in several distinct control modes including (i) "L M1N /L MA χ Mode . and (n) "2'3 Mode, ' and (in) "Full Angle Control Mode " The precise mode in which the control chip 41 is operating determines the type of firing signals that are applied to the power switching devices in the power converter 30 In the L^VL^x Mode the energization of the phase windings is accomplished over an active cycle" that is defined bv predetermined T 0N and T 0FF angles When the L MIN /L M4J< Mode is selected the T 0N angle for a given phase winding coπesponds to the angular position of the rotor with respect to the stator when the inductance of the phase winding is approximately at its minimum value, L MiN In this mode, the T 0FF angle for the phase winding corresponds to the angular position of the rotor when the inductance of the phase winding is roughly at its maximum L MAλ In one embodiment of the present invention each L MIN or point of the three phase windings is associated with a specific output signal from RPT 3 In such an embodiment, the controller (when operating in L MIN /L MAX Mode) initiates the active cycle for a given phase winding when the L MIN signal for that phase windmg is received and terminates the active cycle when the appropriate L MAX signal is received While the T 0N and T 0FF angles for the phase wmdmgs are set and unchanged in the L^/L^ AX Mode, the peak magnitude of the cuπent that is allowed to flow m each phase wmdmg during its active cycle is allowed to vary as a function of the torque demand signal received by the digital core on line 2 Specifically, the digital core 42 mcludes a circuitry for selecting an l^ p ^ ) signal that corresponds to a PWM desired duty cycle defmmg a desired peak current for a given speed/torque combination As those skilled in the art will appreciate, the relationship between torque and current in a reluctance machine is not linear and can vary with the speed of the machine As such, controller 40 includes circuitry that receives the torque demand and speed information and selects an appropriate I,.^,™ ) In one embodiment of controller 40, l ref[pwm) data for vaπous speed and torque points is stored in a sparse matrix stored in memory accessible to the controller 40 In this embodiment the speed/torque information is used to address the matrix memory For speed/torque points for which no stored data is available, interpolation routines may be used to calculate appropriate 1,- f i pwm) data Alternate embodiments are envisioned where the I r - f T pwn , ) duty cycle signal is calculated from speed and torque data or where curve fittmg techniques are used to define an equation for I^ w -,, based on measured data Referring to Figure 4A, the l f -a p ,™, signal generated by the digital core is provided via a pin on the control chip 41 to an external RC filtering network 47 such that an analog signal correspondmg to the desired peak active cycle current Iref (DC) is generated The precise size and structure of the RC network 47 will vary depending on the frequency and magnitude of the 1,-n pw -,, signal on the particular application In one embodiment, the frequency of the I re n pwπ „ signal is approximately 40 Khz, and the RC network 47 consists of resistors 48a-48d and two capacitors 49 where the values of the resistors are, respectively, 2 2 K Ohms, 100 K Ohms, 20 K Ohms and 20 K Ohms, and where the capacitance value of the capacitors 49 is 100 nano- farads The single analog signal representing the desired peak current value is provided via an input pin to comparators 44a 44b and 44c where it is compared with the feedback current signals from the three phase windings A, B and C The comparators 44a, 44b and 44c provide an indication of whether the respective phase cunents are greater than or less than the desired peak cunent reflected bv the Iref (Dπ signal It should be noted that the use of a single pin to receive a single Iref (Dπ signal that is compared to each of the phase winding cunents simplifies the construction and reduces the cost of control chip 41 smce individual cunent references are not required for each of the different phase windings As discussed more fully below, in the L M1N /L MAλ Mode, the control chip 41 receives the cunent compaπson signals and controls the firing signals for the power switching devices such that, when a phase is in its active cycle, the cunent in the phase winding will be maintained at a level approximately equal to the level conesponding to the Iref (DC) signal In one embodiment of the present invention the cunent in the phase windmg during an active cycle is maintained by a fixed frequency soft chopping technique wherein the power switching devices associated with an active phase winding are both actuated into a conductmg state when the cunent in the phase windmg is below the desired peak level reflected by the Iref (DC) and one of the power switches is rendered non-conductive to allow freewheeling when the cunent in the active phase winding exceeds the desired peak cunent level The fixed frequency soft-chopping control scheme discussed above is only one method in which cunent control can be accomplished durmg the active cycle Hard-choppmg techniques can also be used, although such hard choppmg techniques will, for the same number of "chops", result in approximately twice the switchmg losses since two switches must be activated for each chop Alternate forms of soft choppmg may also be used For example, the power switchmg device that is left in a conducting state and the power switching device that is switched between a conductmg and a non-conducting state can be made to alternate to reduce switchmg system losses, extend the service life of the devices, and allow the use of slower switchmg devices The L MrN /L MAX Mode discussed above is most suitable for operation of the reluctance machme 20 at relatively low speeds where the mductance mcreases relatively slowly allowmg a cunent to be established m the wmdmg relatively quickly At high speeds where the mductance changes more rapidly over time, application of fixed duty cycle pulses between L MrN and L MAX may result in a cunent profile in the phase which mcreases and decreases too slowly, thus torque is not easily controlled The L M ^/L MA X mode may be limited for use at relatively low speeds only As an example, in one application of the control system of the present invention, the L MIN /L MΛX Mode is used for rotational speeds up to 375 RPM As the rotational speed of the rotor increases, the negative torque that is generated bv the cunent that remains m an energized phase winding after the L MAX position can result in inefficient motor performance As such, as the rotational speed of the rotor begins to mcrease, it may be desirable to end the active cycle of a given phase wmdmg prior to the L MA * position In the "2/3 Mode", the controller 40 of the present mvention ensures that this occurs by defining the T 0FF angle for each phase wmdmg to be the rotor position that is 2/3 of the angular distance between the L M(N and L MAX positions At this 2/3 point which occurs prior to L MA ^ cunent m the wmdmg can be reduced relatively slowly, while normal forces are low, rather than quickly driven to zero In one embodiment of the system of the present invention this 2/3 point conesponds to a specific RPT output signal In one embodiment, the T 0N angle for an active phase in the 2/3 Mode is the same as that m the Mode ( i e L M ι N ), but it could also be selected independent of the L M1N /L MAX Mode While the T 0N and T 0FF positions are fixed in the 2/3 Mode, the peak desirable cunent is allowed to change as a function of the torque demand signal The method used by controller 40 for setting the peak desirable cunent and indicating whether a given phase winding cunent exceeds this value is that same as that previously described above in connection with the L M1N /L MAX Mode. While L MtN /L MAX Mode is best suited for low speeds where high torque output of the system is an important consideration, the 2/3 Mode of operation is best suited for operation where higher speeds are expected and system efficiency is key. In 2/3 Mode, the angular speed of the rotor is greater than the speed where the negative torque produced using the L MrN /L MAX Mode begins to reduce efficiency of motor performance, but still sufficiently low that there is enough time between the T 0N and T 0FF positions for a sufficient number of cunent chops to be performed such that reliable cunent control is maintained. As an example, in one application of the present invention, the 2/3 Mode is used for rotor speeds above 375 RPM but less than 1000 RPM. In other applications, the 2/3 mode can be used at very low rotor speeds in place of the L Mrf /L MAX mode for better efficiency and lower RMS phase cunent. As the rotational speed of the rotor increases further, a point is reached where the motor inductance and counter emf produced limits the amount of current which can be established in a phase winding using either the 2/3 mode or the L M PVL MAX mode as described above. Further, if a phase winding is de-energized at a T 0FF position conesponding to either L MλX or 2/3 the rotational distance between L MIN and L MA X . lt ' s difficult to remove cunent which may remain in the winding after the point of maximum inductance prior to the next energizing cycle of the winding. Chopping may be done at these speeds, but does not provide adequate cunent control by itself. In this situation, the reluctance machine may be controlled in an "Angle Control Mode" where the energization of the phase winding is defined using different switching schemes at different rotor positions or angles. In one embodiment three angles are used to define discrete +HVDC, freewheeling and -HVDC schemes, although other more complicated embodiments are envisioned, using different angles and more or less than three angles. In this "three-angle" example, the three angles that define the energization of a phase winding being controlled in the Angle Control Mode are: (i) the T 0N angle that defines the initiation of the active cycle; (ii) a freewheeling angle ("FW") where the cunent in the phase winding is allowed to freewheel through one of the power switching devices and one of the fly-back diodes associated with the winding; and (iii) the T OFF angle that defines the termination of the active cycle. Unlike in the L MΠM /L MA X and 2/3 Modes, the T 0N , T 0FF (and FW) angles are not fixed and can vary with the speed and torque demand of the motor in the Angle Control Mode. Moreover, the rotor positions associated with the T 0N , T 0 FF ar, d FW angles often do not conespond to specific RPT signals such that methods must be used to estimate the rotor's position between successive RPT signal changes. Because the T 0N , T 0FF and FW angles vary as functions of the angular speed of the rotor and of the torque or speed demand of the machine, the controller must be provided with (or generate) appropriate angle parameters for each combination of rotor speed and torque that can be encountered while the controller is operating in the Angle Control Mode. One approach to providing such information is to "characterize" the motor by experimentally running the motor in the Angle Control Mode and determining, for several speed/torque points, the appropriate T 0N , T 0FF and FW angles that provide desirable operating performance at the various points. That data can then be stored in a digital memory associated with control 41 (e.g., a memory located on chip 41 or an external memory addressable and accessible to control chip 41 for use by the digital control core. In one embodiment, the selected angle parameters are stored in a sparse matrix stored in memory accessible to the control chip 41. The controller either operates from the angle parameters associated with the speed/torque point most near the actual speed/torque point or uses an interpolation algorithm to select the appropriate angle parameter data for the actual speed/torque operating point Alternate embodiments are envisioned where the T 0N , T 0FF and FW angles are calculated or otherwise derived from the speed and torque information available to controller 40 The precise method that should be used to characterize a motor for operation in the Angle Control Mode will vary from motor to motor and controller to controller General techniques for performing the characterizing function are known in the art and will not be addressed herein in detail Angle Control mode is best suited for high speed machine operation, i e when the machine is operating above a predetermined percentage (e g , between 1/3 and 1/2) of its maximum operating speed As an example, in one application of the present invention Angle Control mode is used for control puφoses when the angular speed of the rotor exceeds 1000 RPM The controller 40 of the present invention may be configured to operate in only one of the operating modes discussed above or to operate in different operating modes depending on the rotational speed of the machme When the latter approach is desired, the controller can be suitably programmed, by stormg the speeds at which a mode change is to occur, in digital memory locations withm the controller 40 As discussed above, abrupt changes m the magnetic flux within the reluctance machine cause unwanted acoustic noise and vibrations The majority of these abrupt changes occur when the cunent in a phase wmdmg is decreasing from its magnitude m the active cycle to near zero (the "back-end' of the cunent waveform), and when the cunent in an active phase windmg is mcreasmg from near zero to its peak value in the active cycle (the "front-end" of the cunent waveform) The mcreasmg cunent can cause changes not only in the flux of the energized wmding, but may also cause changes in the flux of the adjacent wmdmg which is energized or a phase which is being de-energized The controller 40 of the present invention can control the net average voltage applied to the cunent in a phase wmdmg durmg the active cycle in the three previously discussed modes such that the changes in the magnetic flux in an active phase winding or an adjacent phase windmg are reduced, which will reduce unwanted machine noise and vibration This cunent control is accomplished at the front-end of the cunent waveform by "ramping-up" the cunent in a controlled fashion, and at the back-end of the current waveform by "ramping-down" the cunent in a controlled fashion For puψoses of discussion, the method and apparatus used by controller 40 to "ramp-down" the cunent waveform will be discussed first The following discussion is applicable to ail operating modes and is not dependent on the manner in which T 0N and T OFF are defined or selected The simplest way to drive down the cunent in an active phase wmding at the end of the active cycle for that winding is to actuate both power switching devices associated with that winding into a non-conductive state when the rotor reaches the T 0 FF position for that active cycle This technique, however, results in the application of approximately the full -HVDC bus voltage level to the phase winding at T 0FF and results in a dramatic change in the net average voltage applied to the winding from a net positive voltage (whateverthe voltage was that was used to establish the desired active cycle cunent) to -HVDC The abrupt voltage change tends to cause abrupt magnetic flux changes and results in unwanted motor noise and vibration Controller 40 is configured to allow the cunent to * ramp-down" in a predetermined manner such that the cunent changes in the active phase winding are not abrupt, but are controlled from the point at which the rotor reaches the T 0FF angle for the active phase until the cunent in that phase is reduced to near zero In general, this is accomplished bv implementing a switching scheme where, at T 0FF , a duty cycle is calculated or captured such that the net average voltage applied to the phase winding just after T 0FF is the same as it was at T 0FF The duration of the voltage pulses applied to the winding is thereafter controlled such that the net average voltage applied to the winding varies in a controlled fashion from its positive value at T 0FF to zero and then to a negative value until the cunent in the phase winding is reduced to near zero The result of controlling the voltage pulses applied to the winding is to ramp down the cunent and the magnetic flux in a smooth manner, while the result of applying the equivalent net average voltage after T 0FF is to create slight changes in the cunent waveform from the point when cunent starts to be removed from the windmg to the point where the cunent approaches a value near zero This controlled cunent change reduces the audible noise and unwanted vibration in the system In one embodiment controller 40 uses a soft-chopping scheme to maintain the cunent in the phase winding at its desired peak value during the active operating cycle In this embodiment, at T 0FF , controller 40 will initiate a hard- chopping scheme, applying voltage pulses to drive the cunent down to zero in a controlled fashion In one embodiment of the system described herein, the duty cycle of the hard choppmg voltage pulses applied to the winding at T 0FF is half of the soft chopping duty cycle at T 0FF , as a percent of the fixed PWM frequency, plus fifty percent While the following discussion will be limited to a detailed discussion of this embodiment of controller 40, it will be apparent to those of ordinary skill in the art that the devices and techniques discussed herein can be used in conjunction with other choppmg or cunent control schemes Figure 5A illustrates in flowchart form a method 50 m which the control system of the present mvention causes the net average voltage applied to the phase windmg to change in a non-abrupt and controlled manner from approximately the net average voltage bemg applied at the T 0FF point to -HVDC The controller 40 accomplishes this by first determining an initial duty-cycle of hard-choppmg voltage pulses that corresponds to a net average voltage equal to or near the net average voltage applied at the T OFF point This initial duty cycle is thereafter adjusted to control the net average voltage applied to the phase winding, and to "ramp-dowri" the cunent in a de- energized phase wmdmg through a controlled application of hard-chop voltage pulses Prior to the phase windmg entering its active cycle, the digital core 42 of controller 40 calculates the initial duty cycle of the hard-choppmgpulses that will begin at the T 0FF point This is accomplished in steps 51 -53 of Figure 5 A Since the net average voltage required to establish a desired cunent magnitude in a phase windmg vanes with the speed and torque of the machme, for optimum noise reduction, the duty-cycle for which hard-choppingshould begin at the T 0FF point (which defines the net average voltage at the T 0FF point) should vary as a function of both the rotor speed and the machine torque output As such, the controller 40 of the present invention calculates in step 51 a contribution to the "HARD-RAMP-START' duty cycle (l e , a contribution to the net average voltage at T OFF ) as a function of the ratio of the output motor torque to the maximum possible rotor torque In the specific example of Figure 5A, the torque-related contribution to the HARD-RAMP-STAR1 duty cycle is calculated by a processor in the digital core 42 as a linear function of the torque output where HARD-RAMP- START TOR0UE CON TR IB * M -r OFFSET, where M represents a desired slope of a representative linear curve and OFFSET represents a desired offset The precise values of M and OFFSET will vary from machine to machme and may be determined experimentally in a manner similar to the characterizing" process described above where different M and OFFSET values are tested and the optimal M and OFFSET values for a given machine are determined. M and OFFSET may vary with speed and torque and OFFSET can be zero in some applications. Characterization techniques may be beneficial since the optimum relationship between speed, torque, and the desired net average voltage at T OFF (defined by HARD-RAMP- START) is often not linear, but is instead defined by higher order polynomial equations. After calculating the torque contribution to the HARD-RAMP-START duty cycle in step 52, the controller 40 of the present invention calculates the speed contribution to the HARD-RAMP-STARTduty cycle in step 52. This may be done in a manner similar to that previously described above in connection with the torque contribution to the HARD-RAMP-STARTduty cycle. The torque and speed contributions to the HARD-RAMP-START duty cycle are summed in step 53 to produce the desired HARD-RAMP-START value for the given speed and torque. The value of HARD-RAMP-START could alternativelybe calculated using a single function which intrinsicallycombines both factors. Although the above describes methods for calculating the HARD-RAMP-STARTduty cycle through the use of linear functions, other approaches may be adopted. For example, higher-order equations could be used for calculation puφoses, or HARD-RAMP-START data for various speed and torque points could be predetermined and stored in a sparse matrix in the controller 40. As with the angle parameters discussed above, inteφolation routines could be used to calculate HARD- RAMP-START values for torque/speed points not pre-stored in the sparse matrix. While the above examples reflect a HARD-RAMP-STARTduty cycle that varies linearly with the speed and torque of the machine, non-linear relationships may also be used as well as estimation or adaptive control methods. Such alternate schemes, while requiring more computational overhead could provide better results. Figure 5B illustrates several possible HARD-RAMP-START duty cycle curves for various speeds and various torque output levels. These curves were determined experimentally for a given reluctance machine by monitoring the noise and voltage produced by the machine and selecting the HARD-RAMP-STARTduty cycle that, for a given speed and torque point, produced the least unwanted noise. Curves such as those illustrated in Figure 5B could be experimentally obtained and then stored in a memory accessible to the digital core 42 of controller 4, or curve-fitting techniques could be used to determine the equation that most closely fits the experimentally determined data. In the controller 40 of the present invention, at the position T 0FF , hard-chopping at the HARD-RAMP-START duty cycle is initiated, creating a convex portion or "rounded corner" in the current profile at T OFF . By creating this rounded comer, rapid changes in the voltage applied to the winding and the cunent and fluxes induced thereby are avoided, reducing the rate of change of magnetic flux. As previously discussed, rapid changes in the flux are a primary source of acoustic noise and unwanted vibration. The hard-chopping duty cycle is thereafter adjusted to control the net average voltage applied to the phase winding such that it smoothly transitions from the positive value it had at T 0FF to a negative value sufficient to drive the cunent in the phase winding down to zero. Figure 5C illustrates the phase cunent in a phase winding 55, and the voltage across the phase winding 56. Figure 5D illustrates the phase cunent in a winding 55, and the net average voltage across the phase winding 58. In 2/3 mode shown, the controlled ramp down creates a smooth transistion or rounded comer in region 57 at T 0FF . Figure 6A illustrates in flowchart form one method 60 in which the controlled-rampdown is accomplished by the controller 40 of the present invention. Initially at step 50 the controller determines an appropriate HARD- RAMP- START duty cycle for the actual speed and torque parameters This may be accomplished by the method 50 discussed in connection with Figure 5 A or any of the other methods for determining HARD- RAMP-STARTdiscussed above After HARD-RAMP-START is selected, the controller then selects in step 61 a RAMPDOWN-GRADIENT parameter that controls the rate of change of the duty cycle that is used in the ramp-down This RAMPDOWN- GRADIENT parameter controls the rate at which the net average voltage changes from positive to negative across the phase winding and, thus, the rate of change of the cunent in the phase winding and the resulting magnetic flux in the machme The RAMPDOWN-GRADIENTparametermay be calculated as a function of rotor speed and machine torque in a manner similar to that previously described above in connection with the calculation of the HARD-RAMP-START duty cycle As before, the slope and offset values of the linear equation defining RAMPDOWN-GRADIENT as a function or rotor speed and torque can be experimentally determined The other approaches discussed above in connection with the determination of the HARD-RAMP-START value can also be used to determine the RAMPDOWN-GRADIENT Figure 6B illustrates some expeπmentally deπved values for the RAMPDOWN- GRADIENT for a given reluctance machme, as well as some additional linear and higher order curve fits As may be noted the relationship between the optimum RAMPDOWN-GRADIENTand speed and torque is not linear although a linear approximation may be used for simplified control puφoses For the example of Figure 6B, the value of RAMPDOWN-GRADIENT is inversely proportional to the rotational speed of the machine In general, the RAMPDOWN-GRAD should be selected such that the negative slope of the cunent dunng rampdown mcreases as the speed of the machine increases RAMPDOWN-GRAD may also be selected such that the negative slope of the cunent durmg ramp-down vanes as torque increases to improve machme efficiency After the RAMPDOWN-GRADIENT is selected in step 61, the controller 40 then determines whether the rotor has reached the T OFF point for the appropnate phase wmding This may be accomplished by comparing a digital value representing the rotor position (that is derived from the RPT signal) with a digital value representing the T 0FF point that is either fixed for L MIN /L MAX and 2/3 Mode or provided to (or calculated by) the controller for Angle Control Mode Once the controller has determined that the rotor has reached the T 0FF position it then sets a PWM-HARD duty cycle parameterto HARD-RAMP-START and begms applymg power to the appropnate phase wmdmg at a duty-cycle conespondmg to PWM-HARD parameter The controller then enters into a loop where the controller repeatedly adjusts the PWM-HARD duty cycle to reduce the voltage applied to the phase wmdmg as a function of the RAMPDOWN-GRAD in step 65 and returns to step 64 where it begms to apply voltage to the appropriate phase wmdmg at the new and reduced PWM-HARD This cycle continues until the PWM-HARD parameter conesponds to a 0% duty cycle such that the full-HVDC voltage is applied to the phase winding, or until a rotor position is reached where -HVDC is applied In one embodiment of controller 40, the HARD-RAMP-START and RAMPDOWN-GRAD parameters are numbers that are used by various counters and comparators to generate the appropriate firing signals Circuitry conespondingto this embodiment is illustrated in Figure 6C This circuitry may be contained within the digital core 42 of control chip 41 or may be emulated through the use of an appropriately programmed microprocessor or microcontroller Refening to Figure 6C, an 8-bιt comparator 66 receives at its A input the ouφut of an 8-bit up counter 67 and at its B input the ouψut of an 8-bit down counter 68 Comparator 66 provides an ouφut signal (HARD-CHOP- FIRTNG-SIGNAL)that, in the embodiment of Figure 6B, is positive (or logic high) whenever the 8-bit value at A is less than B and that is approximately ground (or logic low) otherwise The HARD-CHOP-FlRING-SlGNALmay then be used by other circuitry (including driver circuitry, not shown) to actuate both power switching devices associated with the relevant phase winding into a conductive state when HARD-CHOP-FIRING-SIGNAL is high and to actuate both power devices into a non-conducting state when HARD-CHOP-FIRING-SIGN ALis low Up-counter 67 is configured such that the counter counts up from 0 to the maximum count value and then resets itself at a rate that conesponds to the frequency of the CLK signal Thus, the output of up-counter 67 and the A input of comparator 66 (refenedto as PWM-COUNT), cycles from 0 to a maximum value at a fixed frequency Down-counter 68 receives at its data input a digital value conespondingto the HARD-RAMP-START value determined by the controller Thus, when enabled by the RAMPDOWN-GRAD as shown below, down-counter 68 will count down from HARD-RAMP-START to zero at a rate defined by the CLK signal The enable signal to down- counter 68 is provided by the count equals zero ouφut of another 8-bit down counter 69 Down-counter 69 is configured to repeatedly count down from its data in value (which is the RAMPDOWN-GRAD) to zero When the counter reaches zero, it generates a CO pulse that is applied to the enable mput of down counter 68 and to the load input of down counter 69 Thus, down-counter 69 will enable down-counter 68 for a smgle clock pulse at a rate mversely proportional to the value of RAMPDOWN-GRAD Thus, the lower the value of RAMPDOWN-GRAD, the more frequently the CO pulses are generated and the faster the down-counter 68 counts down The faster down-counter 68 counts down, the faster the duty cycle of the hard-chopping pulses drops and the faster the net average voltage applied to the phase wmding changes from positive to negative It should be noted that the circuitry of Figure 6C would be repeated for each phase wmdmg. The ramp-down techniques discussed above represent examples of the types of rampdown techniques that may be used with controller 40 Specifically, the rampdown techniques described above were m connection with a hard-choppmg scheme that was initiated at T 0FF It is possible to implement a similar controlled rampdown of the phase current that began usmg a soft-chopping scheme at T 0FF accordmg to Figure 3B(V) which results m the same net average voltage being applied at the T OFF point as would be provided with a hard-chopping scheme chopping at the HARD-RAMP-STARTduty cycle, and then ramping down the soft chopping duty cycle using techniques similar to those descnbed above until the net average voltage applied to the appropriate phase wmdmg is zero, or a desired value above zero The controller could then switch to a hard-choppmg scheme to drive the net average voltage negative or could switch to a soft-chopping scheme, accordmg to Figure 3B(VI) to accomplish the same function As examples, Figure 7A illustrates the phase cunent in a winding 70, and the voltage aplied to the windmg 71 , where a soft-chopping switching strategy is applied at T 0FF 72, the duty cycle is ramped to zero, and then a hard- chopping strategy is employed in region 73 Figure 7B illustrates the phase cunent in a winding 75, and the voltage applied to the winding 76, where a soft-chopping switching strategy which applies the +HVDC voltage and a zero voltage is applied at T 0FF 77, the soft- chopping duty cycle is ramped to zero, and then a soft-chopping switching strategy which applies the -HVDC voltage and a zero voltage is employed in region 78 Different RAMPDOWN-GRADIENT values could be used for the different soft-chop and the hard-chop sections It should be noted that, for a given hard-chopping duty cycle, the same net average voltage will be produced by a soft-chopping duty cycle equal to twice the value of the hard-chop duty cycle minus 50% of the full period of a duty cycle One of ordinary skill in the art having the benefit of this disclosure should be able to implement these alternate ramp down techniques The implementation of the rampdown techniques described above result m a largely convex waveform from the T 0FF point, or a "rounded comer ' which has been found to result in reduced noise The ramp-down feature of controller 40 is beneficial in reducing noise and unwanted vibration for all control modes at all motor torque and speed points At high rotational speeds, however, it may be desirable to drive the cunent in a given phase winding to zero as rapidly as possible following the rotor reaching the T 0FF angle for that phase winding to avoid producmg negative torque As speed increases, RAMPDO WN -GRADIENT may be reduced to zero, effectively disabling the ramp-down functionality at high speeds and allowing the controller to render both power switching devices associated with the phase winding non-conductive at T 0FF , thus applying a full -HVDC voltage across the wmdmg and driving the phase windmg cunent to zero as rapidly as possible In one embodiment of the system of the present mvention, the ramp-down circuitry is disabled at rotor speeds above 1450 RPM In addition to providing for a controlled ramp-down of the cunent waveform, controller 40 also allows for a controlled ramp-up of the phase windmg cunent and efficient fixed frequency chopping control of the phase wmdmg durmg the active cycle once the peak desired cunent is reached The basic method used by controller 40 for ramp-up and peak cunent control is illustrated in flowchart form m Figure 8 A Refenmg to Figure 8A, prior to the initiation of an active cycle, the controller determines a maximum chopping pulse duration for the active cycle that defines the maximum width of a chopping pulse dunng the active cycle as a function of the signal on Ime 2' In one embodiment, this parameter conesponds to a soft-chop pulse width that applies the same net average voltage, or a slightly greater net average voltage, as the duty selected for the HARD- RAMP-STARTduty cycle previously discussed In such an embodiment this PWM-SOFT-MAXduty cycle parameter is calculated to be equal to twice the value of the HARD-RAMP-START duty cycle minus 50% of a fixed frequency period Thιs PWM-SOFT-MAXvalue ιs calculatedιn step 81 The use of a PWM-SOFT-MAX that conesponds to the HARD-RAMP-START is beneficial because it reduces the relative complexity of the system in that it does not require extensive calculations to determine PWM- SOFT-MAX However, m some applications it may be desirable to determine PWM-SOFT-MAX independently of HARD-RAMP-STARTusmg the methods previously discussed for the development of the appropriate HARD-RAMP- START values for different speed and torque combinations While the following discussion is in the context of a PWM-SOFT-MAX value that conesponds to the HARD-RAMP-START value it should be appreciated that the PWM- SOFT-MAX value could be separately determined, and that the HARD-RAMP-START value could be determined from the PWM-SOFT-MAX value After determining the PWM-SOFT-MAX in step 81 the controller determines, for each phase winding whether the T os position for that phase winding has been reached When the position information derived from the RPT 3 indicates that T 0N for the phase winding has been reached the controller 40 generates firing signals to actuate both of the power switching devices associated with the phase winding into a conductive state thus applying the full ^HVDC bus voltage to the relevant phase winding at the frequency and duty cycle conesponding to PWM-SOFT- MAX As set forth more fully below, fixed frequency chopping is implemented through the use of a counter that begins counting and reaches its maximum value at the end of the pulse cycle only to begin counting again The output of this counter refened to a CYCLE-COUNT is used in the controller 40 of the present invention to provide stable, fixed-frequencypulse width modulation Once T 0N is reached and both power switching devices are rendered conductive, the full +HVDC power will be applied to the relevant phase winding until one of two events occurs The controller continuously monitors the ouφut of the cunent comparator 44 for each phase wmding (See Figure 4A) to determine whether the cunent in the relevant phase winding has exceeded the I re(iDC) that defines the maximum desirable peak phase wmdmg cunent If the phase cunent has exceeded the desired maximum value, determined in step 84, the controller will actuate one of the switchmg devices into a non-conductive state, step 85, and allow the cunent to decrease slowly as it freewheels through the conductive power device and the appropriate fly-back diode The power switchmg devices will remam in this freewheeling state until the counter that defines the CYCLE-COUNT counter is reset in step 86 Upon reset of the counter, both power switchmg devices will agam be actuated into a conductive state, step 83, and the control cycle will repeat If the phase cunent is below the maximum desired value, both power switchmg devices will remam conductive until the CYCLE-COUNT exceeds the PWM-SOFT-MAX count When the CYCLE-COUNT is determined to exceed the PWM-SOFT-MAX, step 87, the power switchmg devices are actuated into a freewheeling arrangement, step 85, and remam m that arrangement until the CYCLE-COUNTcounter is reset In controller 40 of the present mvention, immediately after T 0N , when the cunent in the phase wmdmg begms to increase from zero, the PWM-SOFT-MAX count will be reached before the cunent m a given CYCLE-COUNT exceeds the I re( rrχ: ) value Thus, in this ramp-up region, the PWM-SOFT-MAXserves the function of limiting the rate at which the cunent in the phase wmding mcreases Eventually the cunent in the phase windmg will reach a pomt where, for each CYCLE-COUNT, the phase current exceeds the I re( r DC) value before the PWM-SOFT-MAX count is reached Moreover, because the inductance mcreases from or slightly before or after the T ON position is reached, the pomt in the CYCLE-COUNT where the phase winding cunent exceeds I Γ - ITDC) will shift from a value nearer the start of the PWM cycle (when the phase winding mductance is low) to a point near or past PWM-SOFT-MAX as the phase wmdmg mductance approaches its maximum value Thus, m operation, the width of the soft-chopping pulses that are applied to the phase wmding will vary from a duty cycle defined by PWM-SOFT-MAX durmg the ramp-up cycle of the cunent waveform, to a more narrow duty cycle when the phase inductance is low (because of the limiting effect of WD C >) > t0 a duty cycle that is very nearly PWM-SOFT-MAX at the T 0FF point Figure 8B illustrates cunent m a phase wmdmg 88 increasing at a constant PWM-SOFT-MAX duty cycle until the cunent exceeds I ref} iχ> In addition to providing advantages in controlling the RAMP-UP of the phase cunent, the controller of the present invention ' s limiting of the maximum chopping duty cycle during the active cycle is beneficial because it allows for a fixed frequency chopping scheme that does not suffer from cunent programming instability As those of ordinary skill in the art will recognize, fixed frequency converters suffer from a potential cunent programming instability which may result in pulses being missed or skipped, or the duty cycle changing abruptly, such that fixed frequency chopping is not maintained This problem is well recognized and is discussed, for example, in Shu et al "Modeling and Analysis of Switchmg DC to DC Converters and Constant Frequency Cunent Program Mode ' presented at the Power of Electronics Specialist Conference in 1979 Typically complicated analog slope compensation circuitrv is used to address this problem The controller of the present invention solves the cunent programming instability problem through the use of the PWM-SOFT-MAX limiter and the selected switching strategy implemented by controller40 In particular because the chopping pulses in the active region are limited to a specific maximum duty cycle, fixed frequency chopping will always occur Moreover, the use of the particular scheme adopted in controller 40 insures that any variances between the desired cunent and the actual phase cunent will be minimized over time This is accomplished at low speeds by insuring that the PWM-SOFT-MAX value is set at less than 50% of the fixed frequency duty cycle At higher speeds and higher cunents where larger duty cycles are required, instability is avoided through the use of soft chopping m the active cycle This soft choppmg solves the instability problem because the rate of change of cunent increase in a soft choppmg scheme is greater than the rate of change for cunent decrease during freewheeling This difference in the slopes of the cunent increase and cunent decreasing portions solves the instability problems Figure 8C illustrates exemplary circuitry for implementing both the RAMP-UP and RAMP-DOWN control functions discussed above In general, Figure 7B is similar to the circuitry of Figure 6C with the primary exception being the addition of a multiplexor 81 which receives as its input the ouφut of DOWN-COUNTER 68' , and a digital signal correspondmgto the PWM SOFT-MAX value. A control input 82 determines which of the inputs to multiplexor 81 passes through to the B input of comparator 66' Dunng operation, circuitry not illustrated detects whether the phase windmg conespondmgto the circuitry of 8C is m the active cycle (defined by the detection of the rotor reachmg the T 0N position) When the active cycle is detected, the input 82 to multiplexor 81 is set such that the PWM SOFT- MAX value passes through multiplexor 81 to the B input of comparator 66' Thus, from this point the comparator will produce pulses having a width that is defined by the PWM SOFT-MAX value The ouφut of comparator 66' may be provided to circuitry, not illustrated, which converts the ouφut of comparator 66' to soft chopping pulses having a duty cycle defined by the lesser of (I) the point where the phase cunent reaches the digital value conespondmgto l reΛD c> or (n) the PWM-SOFT-MAX value Once the active cycle ends, as reflected by the detection of the T 0FF position, the control input 82 to multiplexor 81 will change such that the digital value provided by down counter 68' is passed through multiplexor 81 to the B input of comparator 66' An appropriate enabling signal may then be applied to DOWN-COUNTER 69' such that DOWN-COUNTER 69' is enabled only after the T 0FF position is detected Upon activation of DOWN-COUNTER 69', the circuitry of Figure 8C will serve to RAMP-DOWN the phase cunent in a manner previously described m connection with Figure 6C While the circuitry of Figure 8C uses the HARD-RAMP- START value to define the choppmg duty cycle or pulse width at the T 0FF point, in alternate embodiments the cycle count at which the chopping pulse just prior to T 0FF terminated is detected, and used in place of the HARD-RAMP START value when RAMP-DOWN begins The detection of the cycle count at the time the last complete chopping pulse in the active cycle occurs may be monitored through simple circuitry and loaded as HARD-RAMP-START prior to T 0FF The use of this circuitry while slightly more complex, insures that there are no changes in the net average voltage applied to the phase winding before and after turn off While the above discussion has been generally directed to fixed frequency hard-choppmg and soft- chopping techniques, those skilled in the art will recognize that other switching strategies could be employed to control the voltage applied to a phase winding. Such switchmg strategies could include fixed on-time or fixed off- time variable frequency techniques, voltage notching techniques, and the like In addition, alternate embodiments are envisioned, and one skilled in the art will recognize, that cunent control techniques could be applied in place of the voltage control strategies employed by the cunent embodiment. Figures 9A and 9B illustrates representative cunent wave forms for a motor operating with and without the RAMP-UP and RAMP-DOWN techniques described above Figure 9A generally illustrates a cunent wave form where the cunent rises dramatically in the region 80 from a zero value to the maximum peak value Chopping is then initiated to control the peak phase cunent until the TURN-OFF point for the phase winding is reached at 82 The power switching devices associated with the phase-winding are then rendered non-conductive and the cunent is driven shaφly down to zero. As a review of Figure 9A indicates, there are abrupt cunent changes at both the front end and tail ends of the cunent wave-form which tend to produce unwanted noise and vibration. Figure 9B reflects a cunent wave-form for which the RAMP-UP and RAMP-DOWN techniques of the present invention were applied after the T 0N point in the region 80' The cunent does not rapidly increase but mstead increases m a controlled manner to the maximum peak cunent value. Also, after the T 0FF pomt 82' the tail cunent does not drop off suddenly but is mstead ramped down slowly such that abrupt changes in the applied voltage, the cunent in the phase winding, and in the magnetic flux are reduced, resulting in a reduction of unwanted noise and vibration. Figure 9B also illustrates the cunent tail portions 84 corresponding to one RAMPDOWN-GRADIENT value. The above description and several embodiments of the present invention are made by way of example and not for puφoses of limitation. Many variations may be made to the embodiments disclosed herein without departing from the scope and spirit of the present invention. For example, while the above description is directed to a specific switched reluctance motor system and control device, the present invention is applicable to any form of reluctance machine regardless of the number of poles, pole shape and general layout and to machine systems that include controllers constructed through discrete digital components or analog circuits. The present invention is intended to be limited only by the scope and spirit of the following claims.

Claims: CLAIMS WHAT IS CLAIMED IS: 1 A method for controlling the rate of cunent change in a phase winding of a reluctance machine, the phase winding being coupled by a power switching device to a source of DC voltage, the method comprising the steps of selecting a maximum net average voltage for application to the phase winding the maximum net average voltage being less than available DC voltage, and apply ing the maximum net average voltage to the phase winding when the rotor reaches a predetermined angular position with respectto the stator 2 The method of claim I wherein the maximum net average voltage is selected as a function of the angular speed of the rotor 3 The method of claim 2 wherein the maximum net average voltage increases with increases in the angular speed of the rotor in a non-linear manner 4 The method of claim 1 further compπsing the steps of receiving an indication of the desired torque output of the machine, and selecting the maximum net average voltage as a function of the indication of the desired torque ouφut of the machine 5 The method of claim 1 further compπsingthe steps of monitoring the cunent in the phase winding and comparing a representation of the cunent in the phase winding with a representation of the desired peak phase cunent, applying the maximum net average voltage to the phase winding from the point at which the rotor reaches the first angular position until the point where the representation of the phase cunent reaches the representation of the desired peak phase cunent, and for a period thereafter applying the lesser of (i) the net average voltage required to keep the phase cunent at a level approximately equal to that of the desired peak cunent or (n) the maximum net average voltage 6 The method of claim 1 wherein the step of applying the maximum net average voltage to the phase winding when the rotor reaches a predeterminedangular position with respectto the stator is accomplished by applymg a pulse width modulated voltage signal to the phase winding, where the peak voltage magnitude of the pulse width modulated signal is approximately the magnitude of the available DC voltage and where the duty cycle of the pulse width modulated voltage conesponds to the desired net average voltage 7 A method of controlling the cunent in the phase winding of an electric machine through the use of fixed frequency chopping, the method compπsingthe steps of establishing a fixed chopping frequency for applying voltage pulses to the motor where the interval between the initiation of adjacent voltage pulses defines a fixed frequency duty cycle, selecting a desired maximum duty cycle, where the desired maximum duty cycle is less than the fixed frequency duty cycle. apply ing voltage to the phase winding at the initiation of a fixed frequency duty cycle, removing the application of voltage from the phase winding at the earlier of ( i ) the phase cunent reaching a desired peak current value, or ( n) the expiration of the desired maximum dutv cycle 8 The method of claim 6 wherein the electric machine includes a rotor and the desired maximum duty cycle is selected as a function of the angular speed of the rotor 9 The method of claim 7 wherein the apparatus receives an indication of the desired torque output of the machine and wherein the desired maximum duty cycle is selected as a function of the desired torque output of the machine 10 The method of claim 6 wherein the desired maximum duty cycle is less than 50% of the fixed frequency duty cycle 1 1 The method of claim 6 wherein the electric machine is a switched reluctance machine wherein the phase winding is coupled across a DC bus by power switching devices, and wherein the step of removing the application of voltage from the phase winding comprises the step of actuating the power switching devices to place the phase winding m a freewheeling condition 12 The method of claim 6 wherein the electric machine is a switched reluctance machine, wherein the phase winding is coupled across a DC bus by power switching devices and flyback diodes, wherein the step of applying voltage to the phase winding comprises the step of actuating the power switching devices such that positive DC bus voltage is applied to the phase winding and wherem the step of removing the application of voltage from the phase winding compπses the step of actuating the power switching such that the negative of the DC bus voltage is applied to the phase winding via the flyback diodes 13 A method of reducing noise in a switched reluctance machine system including a reluctance machme having a rotor and a phase winding, the method comprising the step of controlling the rate of change of magnetic flux in the machine during an interval when the phase cunent rises from a value of approximately zero amps to a desired peak cunent value 14 The method of claim 13 wherein the rate of change of the magnetic flux in the machine is controlled by controlling the net average voltage applied to the phase winding during the defined interval. 15 The method of claim 14 wherein the net average voltage applied during the defined interval is selected as a function of the speed of the machine 16 The method of claim 14 wherein the net average voltage applied during the defined interval is selected as a function of the desired output torque of the machine 17 The method of claim 13 wherem the rate of change of the magnetic flux in the machine is controlled by control ling the rate of change in the cunent in the phase winding during the defined interval 18 A method of enhancing the performance of a reluctance machine system, the system including a rotor and a phase winding coupled to a DC bus via power switching devices, the method compπsingthe steps of (a) controlling the cunent m the phase winding during an interval defined by a first angular rotor position and a second angular rotor position, (b) applying a selected net positive voltage to the phase winding at the point defined by the second angular rotor position. (c) controlling the net average voltage applied to the phase winding as a function of time such that the rate of change the net average voltage applied to the phase winding transitions in a controlled fashion from the selected net positive voltage applied at the point defined by the second angular rotor position to a pre-established voltage value 19 The method of claim 18 wherein the pre-established voltage value is the negative of the DC bus 20 The method of claim 18 wherein the step of controlling the net average voltage applied to the phase winding comprises the step of alternately applying the positive DC bus voltage and the negative DC bus voltage to the phase winding 21 The method of claim 18 wherein the step of controlling the net average voltage applied to the phase winding comprises the step of alternately (i) applying the positive DC bus voltage to the phase winding and ( n) placing the phase winding in a freewheelmganangement 22 The method of claim 18 wherein the step of controlling the net average voltage applied to the phase winding compπses the step of alternately (i) applying the negative DC bus voltage to the phase winding and (n) placing the phase winding in a freewheelmganangement 23 The method of claim 18 wherein the selected net positive voltage is selected as a function of the desired torque output of the machine 24 A method of enhancing the performance of a reluctance machine system, the machine system including a rotor, a phase winding coupled to a voltage bus by power switching devices, the method comprising the steps of (a) selecting a selected net average voltage level as a function of the operating parameters of the reluctance machine, (b) applying the selected net average voltage to the phase winding when the rotor reaches a first pre- established angular position, and thereafter (c) monitoring the cunent in the phase winding and, once the phase cunent exceeds a desired peak cunent value, controlling the cunent in the phase winding to be near the desired peak cunent value, (d) applying the selected net average voltage to the phase winding when the rotor reaches a second pre- established angular position, and thereafter, (e) controlling the net average voltage applied to the phase winding as a function of time such that the net average voltage varies from the selected net average value to a pre-established voltage level in a controlled fashion 25 The method of claim 24 wherem the pre-establishedvoltage level is the negative of the voltage bus 26 The method of claim 24 wherein the net average voltage applied to the phase winding vanes from the selected net average voltage to the pre-establishedvoltage level as a linear function of time 27 The method of claim 24 where the selected net average voltage level is selected as a function of the speed of the machine 28 A method of controlling the energization of a phase winding in a reluctance machine, the machine having a rotor and at least one phase winding, the method comprising the steps of (a) controlling the voltage applied to the phase winding during a first voltage-control interval that begins at a point defined by a first pre-determined angular rotor position and ends when the cunent in the phase winding first exceeds a desired peak cunent value, and thereafter (b) controlling the cunent in the phase winding to be near a desired peak cunent value during a cunent- control interval defined by the point where the cunent in the phase winding first exceeds the desired peak cunent value and ends when the rotor reaches a second pre-determinedangular position, and thereafter (c) controlling the voltage applied to the phase winding during a second voltage-controlled interval that begins when the rotor reaches the second pre-determined angular position and ends when the net average voltage applied to the phase winding reaches a pre-establishedvoltage level 29 The method of claim 28 wherein the net average voltage applied to the phase winding is a constant selected net average \ oltage level 30 The method of claim 29 wherein the net average voltage applied to the phase winding at the initiation of the second vokage-controlledinterval is equal to the selected net average voltage level 31 The method of claim 30 wherein the phase winding is coupled to a DC bus by power sw itching device, wherein the pre-establishedvoltage level is the negative of the DC bus. and the net average voltage applied to the phase winding duπng the second voltage-controlled interval vanes linearly over time from the selected net average voltage level to the negative of the DC bus

Inventors: FRENCH ALAN P, FRENCH, ALAN, P.
Applicants: EMERSON ELECTRIC CO, EMERSON ELECTRIC CO.
Application year: 1997
Main classification: H02P5
IPC codes: H02P6, H02P9, H02P25