RAIL-TO-RAIL INPUT STAGES WITH CONSTANT gm AND CONSTANT COMMON-MODE OUTPUT CURRENTS

Abstract: A rail-to-rail input stage with constant transconductance and constant common-mode output currents is provided. The common-mode output currents are controlled by the use of current switches (M5-M8). When low and high common-mode input voltages are applied to the input stage, the current switches (M5-M8) take part of the tail currents of the input stage transistors (M1-M4), divide it into two equal parts, and direct the two equal current signals to the outputs. When intermediate common-mode input voltages are applied, the current switches (M5-M8) regulate the tail currents of the input pairs such that the common-mode output current and the transconductance remain constant. The current switches (M5-M8) are responsive to changes in the common-mode input voltages, and thereby maintain the transconductance constant.

Description: Rail-to-rail input stages with constant g m and constant common-mode output currents. BACKGROUND OF THE INVENTION Field of the Invention The present invention relates to operational amplifiers. More particularly, it relates to rail-to-rail input stages with constant transconductance and constant common- mode output currents. The Prior Art A rail-to-rail input stage can be composed of an N-type and a P-type transistor in parallel. When the common-mode input voltage of such an input stage moves from one part of the common-mode input range into another, the transconductance changes with a factor of 2. This impedes optimal frequency compensation, since the unity-gain frequency of an amplifier is proportional to the transconductance of the input stage. To obtain an optimal frequency compensation over the common-mode input range, the transconductance of the input stage has to be constant. (J.H. Huijsing and D. Linebarger, "Low- Voltage operational amplifier with rail-to-rail input and output ranges". IEEE J. Solid- State Circuits, vol. SC-20, pp. 1144-1 150, Dec. 1985). Several other methods for controlling the transconductance have been reported for bipolar, as well as CMOS technology. For a complementary input stage designed in bipolar, a current switch is used to obtain a constant transconductance. (J.H. Huijsing and D. Linebarger, "Low-Voltage operational amplifier with rail-to-rail input and output ranges". IEEE J. Solid-State Circuits, vol. SC-20, pp. 1 144-1 150, Dec. 1985). U.S. Patent No. 4,555,673 to Huijsing et al.. discloses a differential amplifier with rail-to-rail input capability and controlled transconductance. The method employed for controlling the transconductance uses current control or current switches to steer at least part of the supply current away from at least one of the differential portions of the input stage when the common mode voltage is in at least one part of the supply range. If the CMOS complementary input stage is operating in strong inversion, a constant transconductance can be obtained by using square root circuits or three times current mirrors. (Botma, J.H., et al., "A low-voltage CMOS operational amplifier with rail- to-rail constant-g,,, input stage and class-AB rail-to-rail output stage" . Proceeding ISCAS93, pp.1314-1317; Hogervorst, R. et al. , "CMOS low-voltage operational amplifiers with constant-g m rail-to-rail input stage". Proceedings ISCAS92, pp.2876-2879; R. Hogervorst, J.P.Tero, R.G.H. Rschauzier, J.H. Huijsing, "A compact power-efficient rail-to-rail input/output amplifier for VLSI cell libraries" , in Digest ISSCC94, Feb 1994.) For CMOS complementary input stages operating in weak inversion a current switch can be used to obtain a constant transconductance (M.D. Pardoen, M.G. DeGrauwe, "A rail-to-rail input/output CMOS power amplifier". IEEE J. Solid-State Circuits, vol. SC-25, pp. 501-504, Dec. 1990). The above described complementary input stages have a common drawback, (i.e. , the common-mode output currents of the input pairs change). These currents load the summing circuit, and vary as a function of the common-mode input voltage. As a result, the biasing currents in the summing circuit vary. For several different reasons this variation is unacceptable. An operational amplifier might need a floating current source, which is, in general, difficult to design (R. Hogervorst, J.P.Tero, R.G.H. Rschauzier, J.H. Huijsing, "A compact power-efficient rail-to-rail input/output amplifier for VLSI cell libraries" in Digest ISSCC94, Feb 1994). The offset contribution of the summing circuit to the input of the operational amplifier changes, resulting in a lowering of the common-mode rejection ratio. To cope with the varying bias currents in the summing circuit, multipath frequency compensation can become very complicated, because each time a path is added, the amount of input transistors has to be doubled. Particularly for rail-to-rail input stages, this results in a relatively large area consumption. The present invention overcomes the drawbacks of previous solutions and provides rail-to-rail inputs stages with constant transconductance and constant common-mode output currents. SUMMARY OF THE INVENTION It is therefore an object of the present invention to provide a circuit for maintaining the common-mode output currents and the transconductance of the rail-to-rail input stage of an operational amplifier at a constant value. It is another object of the invention to provide a circuit for maintaining the common-mode output currents of the rail-to-rail input stage of an operational amplifier constant to facilitate the biasing of the summing circuit connected to the input stage. It is yet another object of the invention to provide a circuit for maintaining the common-mode output currents of the rail-to-rail input stage of an operational amplifier constant such that the summing circuit connected to the input stage does not contribute to offset changes of the amplifier. A further object of the invention is to provide a circuit for maintaining th common-mode output currents of the rail-to-rail input stage of an operational amplifier constant and thereby increase the common-mode rejection ratio of the amplifier. Another object of the invention is to provide a circuit for maintaining the common-mode output currents of the rail-to-rail input stage of an operational amplifier constant to simplify the realization of multipath compensated opamps, and therefore requires less silicon die area. It is a further object of the invention to provide a circuit for maintaining the common-mode output currents of the rail-to-rail input stage of an operational amplifier constant that operates efficiently and reliably. The invention provides a rail-to-rail input stage with constant transconductance and constant common-mode output currents. A current switch is coupled t each of the input pairs of transistors and a voltage source. The voltage source provides reference and bias signals to the gates of the current switches. When common-mode input voltages are applied to the input stage, the current switches compare the input voltage with the value of the voltage source, and determine which of the input pairs should operate. Upon activation of an input pair, the current switches take the tail current of the non-active pair, divide it into two equal parts, and direct the two equal current signals to the output. Thus, the output currents received by the summing circuit connected to the input stage are constant. This control circuit can be embodied using MOS, CMOS, and Bipolar technology. In another embodiment of the invention, the voltage source, connected to the current switches, applies a voltage difference to the gates of the current switches. The application of a voltage difference to the gates of the current switches enables the transconductance values of the current switches to be small with respect to the values of the input pairs of transistors. By decreasing the transconductance values of the current switches with respect to the input pairs, the noise and offset contributions to the input stage caused by the addition of the current switches is significantly decreased, and almost eliminated. BRIEF DESCRIPTION OF THE DRAWINGS Other objects and features of the present invention will become apparent from the following detailed description considered in connection with the accompanying drawings which disclose several embodiments of the present invention. It should be understood, however, that the drawings are designed for the purpose of illustration only and not as a definition of the limits of the invention. In the drawings, wherein similar reference characters denote similar elements throughout the several views: FIG. 1 is a schematic diagram of the input stage of an operational amplifier of the prior art; FIG. 2 is a schematic diagram of a rail-to-rail input stage having a constant common-mode output currents according to the invention; FIG. 3 is a schematic diagram of a constant transconductance rail-to-rail input stage having constant common-mode output currents according to the invention; FIG. 4 is graphical representation of the normalized transconductance versus the common-mode input voltage for the circuit of FIG. 3; FIG. 5a is a BiCMOS embodiment of a constant transconductance rail-to- rail input stage having constant common-mode output currents according to the invention; FIG. 5b is an alternative bipolar embodiment of the constant transconductance rail-to-rail input stage having constant common-mode output currents according to the invention; FIG. 6 is schematic diagram of a CMOS constant transconductance rail- to-rail output stage operating in strong inversion and having constant common-mode output currents according to the invention; FIG. 7 is a graphical representation of the normalized transconductance versus the common-mode input voltage for a CMOS embodiment of the circuit of FIG. 6 with the input stage operating in strong inversion; FIG. 8 is a graphical representation of the normalized transconductance versus the common-mode input voltage for the circuit of FIG. 6 with the input pairs operating in strong inversion; FIG. 9 is a schematic diagram of a CMOS rail-to-rail input stage operating in strong inversion having a constant transconductance and constant common-mode output currents according to the invention; FIG. 10 is a graphical representation of the normalized transconductance versus the common-mode input voltage for the circuit of FIG. 9. ; FIG. 1 1 is a schematic diagram of another embodiment of a rail-to-rail input stage with constant transconductance having constant common-mode output currents according to the invention; FIG. 12 is a schematic diagram of a first alternative embodiment of the circuit of FIG. 1 1 according to the invention; and FIG. 13 is a schematic diagram of a second alternative embodiment of th circuit of FIG. 1 1 according to the invention. DETAILED DESCRIPTION OF PREFERRED EMBODIMENTS FIG. 1 shows an example of an input stage of an operational amplifier of the prior art. As shown, the input stage has a rail-to-rail common-mode input range which can be composed of two complementary differentia! input pairs. The input stage consists of an N-channel input pair, M 1-M2, and a P-channel input pair, M3-M4. The common-mode input voltage range for such an input stage can be divided into three parts: (1) Low common mode input voltages (only the P-channel input pair operates); (2) Intermediate common-mode input voltages; (both the P-channel and N-channel input pair operate); and, (3) High common-mode input voltages (only the N-channel input pair operates). When the common-mode input voltage moves from one part of the common-mode input range into another, the transconductance changes by a factor of two. Thus, since the unity-gain frequency of an amplifier is proportional to the transconductance of the input stage, changes in the transconductance impedes an optimal frequency compensation. Therefore, in order to obtain optimal frequency compensation over the whole common-mode input range, the transconductance of the input stage has to be constant. FIG. 2 shows the complementary input stage, M,-M 4 , with the addition o an output current control circuit 22. Control circuit 22 is a pair of complementary current switches, M 7 -M 8 and M,-M 6 . The gate of the N-channel current switches M 7 -M 8 is biased at a value of V ss +V ref , while the gate of the P-channel switches M,-M 6 is biased at a value of Vss + V^+V*,. The current switches M 5 -M s compare the common-mode input voltage with their respective gate voltages and decide which input pair should be active. For common-mode input voltages above V SS H-V,, r the N-channel input pair M r M 2 is active, whil for common-mode input voltages below V ss +V rcf +V dlf the P-channel input pair M 3 -M 4 is active. When an input pair is active, it splits its tail current into two equal common-mode currents and directs those currents to the output. If the input pair is not active, the current switches divide the tail-current of the non-active input pair into two equal parts which are then directed to the output. As a consequence, the common-mode output currents of the input stage are constant over the entire common-mode input range. The input stage loads the summing circuit with common-mode currents. As a result, the bias currents in the summing circuit do not vary. Therefore, the summing circuit does not contribute to offset changes when the common-mode input voltage swings from rail-to-rail. This results in an increase of the common mode rejection ratio of the input stage. It should be noted, that mismatches of each pair of current switches M 5 -M 8 contributes to the offset change of the amplifier. This contribution can be made small by making the value of the current switches M s -M 8 much smaller than that of the input transistors M,-M 4 . The constant bias currents in the summing circuit also allow a simple realization of multipath compensated amplifiers. FIG. 3 shows a bipolar embodiment of the circuit of FIG. 2. Circuit 30 has a control circuit 32 similar to that of control circuit 22 (FIG. 2), with the difference being that the voltage source v dlf has been made equal to zero. As such, not only the common-mode output currents of the input stage have a constant value, but also the transconductance is regulated at a constant value. Although the circuit is drawn using bipolar transistors, it can also be used for CMOS input stages operating in weak inversion. The current switches, Q 5 -Q 7 keep the sum of the tail-currents of the input pairs C -Q 4 constant. Since, the transconductance of a bipolar transistor is proportional to its tail current, the transconductance of the rail-to-rail input stage, Q 1 -Q 4 , will also be constant. The collectors of transistors Qs-Q 6 can b e connected to the negative supply, while the collectors of transistors Q 7 -Q 8 can be connected to the positive supply V c . Although this results in non-constant common-mode output currents, the transconductance is still controlled constant. To understand the operation of this circuit, the common-mode input range of this circuit can be divided into three parts: 1) If low common-mode input voltages are applied, only the PNP input pair Q 3 -Q 4 operates. The PNP current switches QrQ δ of control circuit 32 are off while the NPN current switches Q 7 -Q κ conduct. As a result, the PNP input pair Q Q 4 is biased by the tail current I rl , which has a value of I rιf . The PNP input pair Q 3 -Q 4 splits its tail current I rl into two equal common-mode parts, and directs those currents to the output of the input stage. The NPN current switches Q 7 -Q 8 take away the tail current I r3 , which also has a valu of I ref , from the non-operating NPN input pair QrQ 2 . The NPN current switches Q 7 -Q 8 also divides the tail current I r2 into two equal common-mode parts and directs these currents to the output. 2) If intermediate common-mode input voltages are applied the PNP as well as the NPN input pairs QrQ 4 operate, and the current switches Q 5 -Q 8 will take away a part of the tail-currents, I rl and I r2 . If the common-mode input voltages increases, the curren through the current switches, Qs-Q 0 , also increases. However, the current through the other pair of current switches, Q 7 -Q 8 , decreases with the same amount. As a result, the sum of th tail currents I r] -I r2 is kept at value of \ rc( . The common-mode output currents also do not change, because the tail currents I r ,-I r2 are directed to the output, through either a current switch Q 5 -Q 6 ; Q 7 -Q 8 or an input pair Q,-Q 2 ; Q 3 -Q - 3) If high common-mode input voltages are applied, only the NPN input pair Q,-Q 2 operates. The PNP current switches Q 5 -Q 6 conduct while the NPN current- switches Q Qg are off. As a result, the NPN input pair Q,-Q 2 is biased by the tail current I r2 , which has a value of I rcf . The NPN input pair Q Q 2 splits the tail current I r2 into two equal common-mode currents, and directs those currents to the output. The PNP current switches Q 5 -Q 6 take away the current I r2 from the non-operating PNP input pair Q 3 -Q 4 , divides it into two equal parts, and directs these currents to the output. Thus, it can be concluded that the common-mode output currents of the input stage are constant. Also, the sum of the tail currents I rl - I r2 of the input pairs QrQ 4 , and therefore the transconductance of the rail-to-rail input stage, is regulated at a constant value. FIG. 4 shows a graphical representation of the normalized transconductance gm versus the common-mode input voltage for the circuit of FIG. 3. As can be seen from the graph, the transconductance gm is approximately constant over the entire common-mode input range. FIG. 5a shows a BiCMOS embodiment 50 similar to circuit 30 shown in FIG. 3. The main difference between the two circuits 50 and 30 is in the noise and offset contribution of the current switches Q . cQx- In FIG. 3, the current switches Qj-Q s split the part of the tail current which is not used for signal amplification into two equal parts. Since these current switches Q 5 -Q 8 are bipolars, they have a transconductance which is comparable to that of the input transistors QrQ 4 . As such, the current switches Qs-Q* contribute to the noise and offset of the input stage. This problem can be overcome in BiCMOS, as is shown in FIG. 5a. In FIG. 5a, the current through the current switches Q Qβ of control circuit 52 is split by the MOS transistors M r M 4 . In this way, the MOS transistors M,-M 4 , instead of the bipolar current switches Q 5 -Q 8 , contribute to the noise and offset of the input stage. This contribution can be made negligible by giving the MOS transistors M,-M a much smaller transconductance than the bipolar current switches Q 5 -Q , which is easily accomplished. This same technique can be applied in CMOS technology, however the additional MOS transistors M,-M 2 should have a much smaller W/L ratio than the current-switches Qs-Q 8 . FIG. 5b shows an alternative bipolar embodiment of the circuit shown in FIG. 3. This circuit includes the rail-to-rail input stage QrQ 4 and current switches Q 5 -Q 8 . The current switches Q 5 -Q 8 are degenerated by the resistors <rR 8 . These resistors R 5 -R g effectively lower the transconductance of the current switches Q .s -Q 8 , and as such, they reduce the noise and offset contributions of the current switches Q s -Q 8 . In order to obtain a constant transconductance, the voltage source VLS should have a value of 0.5*I a . f *R 5 . FIG. 6 shows a CMOS embodiment 60 of circuit 30 of FIG. 3. As shown in the bipolar case of FIG. 3, the voltage source v Jlf is zero. First, suppose the W/L ratios of transistors M 5 -M 6 and transistors M 7 -M„, are equal to the W/L ratio of the input transistors M,-M 4 which are biased in weak inversion. This circuit now operates like its bipolar counterpart (FIG. 3), and the transconductance of the input stage is constant over the common-mode input range. If the input stage operates in strong inversion, the transconductance of the input transistors M,-M 4 is proportional to the square- root of the W/L ratio of the input transistors M,-M 4 . Because the transconductance control keeps the sum of the tail currents of the input pairs M r M 4 constant, the transconductance gm varies about 40% over the entire common-mode input range (FIG. 7). The peak value of the transconductance gm occurs when the common-mode input voltage is equal to the bias voltage of the current switches M 5 - M 8 . At this input voltage the tail-currents of the input pairs M,-M 4 are equal to .5L. f . The input stage can be i proved for operation in strong inversion by making the W/L ratio of transistors M 7 -M 8 and transistors M<;-M 6 three times larger than the input transistor, M,-M 4 . The input stage basically works in the same way as the circuit of FIG. 3, however, in the intermediate part of the common-mode input range, the circuit operates slightly different. If the common-mode input voltage is equal to the bias voltage of the current switches M s -M 8 , the current through the current switches M 5 -M 8 is three times larger than the currents of the input pairs M,-M 4 . As a result, the tail current of the input pairs M r M 4 is equal to .25I rcf , resulting in the same transconductance gm as in the outer parts of the common-mode input range. (See FIG. 8). FIG. 8 shows the normalized transconductance gm versus the common- mode input voltage range for the circuit of FIG. 6 operating in strong inversion. The curre switches M 5 -M 8 have a W/L ratio which is three times larger than the input transistors M,- M 4 (FIG. 6). From this graph, it can be concluded the transconductance gm varies about 17% over the entire common-mode input voltage range. FIG. 9 shows a circuit 90 where the same result can be obtained by applying a voltage difference between the gates of the current switches M 5 -M 8 within control circuit 92. The bias voltage source and the voltage source v Jlf are realized by transistor M, and transistor M l0 . Thus, by giving these transistors -M lu different aspect ratios, a voltage difference between the gates of the current switches M 5 -M 6 and current switches M 7 - M 8 , is created. The main advantage of this approach is that the values of current-switches M 5 -M 6 and the current switches M 7 -M 8 , can be made small compared to the input transistors M,-M 4 , which tends to decrease the noise and offset contribution of the current switches M 5 - M 8 . Moreover, the offset voltage change is spread out over a large part of the common- mode input range which increases the common-mode rejection ratio. FIG. 10 shows the normalized transconductance gm versus the common-mode input range for the circuit of FI 9. From figure 10, it can be concluded that the transconductance gm varies approximately 19% over the entire common-mode input range. FIG. 1 1 shows a rail-to-rail input stage having constant transconductance and constant common-mode output currents for ultipath compensated amplifiers. The circuit consists of a rail-to-rail input stage, QrQ M having constant common-mode output currents, and a summing circuit Q y -Q |6 with multiple outputs. The multiple outputs can be used for multipath frequency compensation of an operational amplifier. The gain path is used for the low-frequency path with high gain, while the multipaths can be used for the lo gain high-frequency path. The signal of the input stage Q,-Q 8 is tapped from transistor Q n . Because the input stage Q|-Q 8 has constant common-mode output currents, the current of th transistors Q 15 -Q 16 al o have a constant value. Therefore, no additional input pairs are needed. In some applications the extra noise and offset contribution of the current switches is not acceptable. In these cases, the variants as shown in FIG. 12 can be used. I this circuit, only the multipath transistors M, 7 -M 20 are biased at a constant current level. This has the same advantage as the circuit of FIG. 1 1 (i.e. , no additional input pairs are necessary). Furthermore, because the control transistors Q s -Q 8 are not connected with the main signal path, there is no additional noise and offset due to the control transistors Q 5 -Q 8 . Note that the summing circuit M 9 -M 14 is biased with a floating current source I r3 . FIG. 13 shows an application which does not require a floating current source to bias the circuit. Although the current levels in the multipath transistors change, the sum of the drain currents of transistor M, 0 and transistor M 17 (i.e. , the drain current of transistor M 21 ) change by the same amount. As a result the multipaths do not have an output current which depends on the common-mode input voltage, and therefore no additional input pairs are necessary. Furthermore, it has the same advantage as the circuit shown in FIG. 12 (i.e., there is no additional noise and offset due to the control transistors or current switches While several embodiments of the present invention have been shown and described, it is to be understood that many changes and modifications may be made thereunto without departing from the spirit and scope of the invention as defined in the appended claims.

Claims: CLAIMS: 1. A circuit for controlling the transconductance (gm) and common mode output currents of the rail-to-rail input stage of an operational amplifier (opamp), the input stage having an output, and at least an N-type and a P-type input differential pair of transistors (M,-M 4 ) each having a gate voltage and a tail current, the opamp having a common-mode input voltage range including low, intermediate, and high common-mode input voltages, the circuit comprising: a first current switch (M 7 -M 8 ) coupled to the N-type input differential pair of transistors (M,-M 2 ), said first current switch (M 7 -M ) having a gate; a second current switch (M 5 -M 8 ) coupled to the P-type input differential pair o transistors (M 3 -M 4 ) said second current switch (M,-M 6 ) having a gate; voltage means coupled to said current switches (M 5 -M 8 ) for providing referenc and bias voltages to the gates of said current switches (M 5 -M ); and, said current switches (M 5 -M 8 ) comparing the common-mode input voltage of t differential input pairs (M,-M 4 ) with the current switch gate voltages, and thereby activates least one of the differential input pairs (M,-M 4 ). 2. The circuit according to claim 1 , wherein said current switches (M 5 -M 8 ) are responsive to changing common-mode input voltages in order to maintain a constant common-mode output current. 3. The circuit according to claim 2, wherein when low and high common- mode input voltages are applied to the input stage, said current switches (M 5 -M 8 ) divide the tail current of a non-active differential input pair into two equal current signals and directs these currents to the output of the input stage. 4. The circuit according to claim 3, wherein when intermediate common- mode input voltages are applied to the input stage, said current switches (M 5 -M 8 ) regulate th tail currents of the active input pairs and direct these current signals to the output of the input stage. 5. The circuit according to claim 4, wherein said current switches (M 5 -M g ) and the input pairs (M,-M 4 ) are a combination of bipolar and CMOS transistors. 6. The circuit according to claim 5, further comprising: additional MOS transistors (M,-M 4 ) coupled to the current switches (Q 5 -Q 8 ); voltage means coupled to the gates of the additional MOS transistors (M,-M 4 ) for providing reference and bias voltage signals to said additional MOS transistors (M,-M 4 ); and said additional MOS transistors (M,-M 4 ) eliminating noise and offset contributions in the input stage caused by said current switches (QrQs)- 7. The circuit according to claim 4, wherein said voltage means applies a voltage difference between the gates of the first (M 7 -M 8 ) and second (M 5 -M 0 ) current switches. 8. The circuit according to claim 7, wherein the application of a voltage difference between the gates of the current switches (M 5 -M 8 ) enables the values of said current switches (M 5 -M 8 ) to be small with respect to the input transistors (M,-M 4 ), and thereby decreases noise and offset contributions to the input stage by said current switches (M 5 -M 8 ). 9. The circuit according to claim 4, further comprising: resistors (R 5 -R 8 ) coupled between the current switches (Q s -Qx) and the input differential pair of transistors (Q,-Q 4 ); said resistors (R 5 -R 8 ) reducing noise and offset contributions of the current switches (Q 5 -Q 8 ) by lowering the transconductance (gm) of said current switches (Qj-Qg)-

Inventors: HOGERVORST RONALD, HUIJSING JOHAN HENDRIK, HOGERVORST, RONALD, HUIJSING, JOHAN, HENDRIK
Applicants: PHILIPS ELECTRONICS NV, PHILIPS NORDEN AB, PHILIPS ELECTRONICS N.V.
Application year: 1997
Main classification: H03F3
IPC codes: H03F3