A system for driving a reactive load

Abstract: The system comprises means (3, G1, G2, 12) for the controlled supply of the reactive load (Cl), for supplying variable quantities of energy to the load in a predetermined manner, reactive components (Cr, Lr) which are connected to the load (Cl) by means of a controllable electronic switch (T3) and which form a resonant circuit with the load when the electronic switch (T3) is closed, means (10) for activating the electronic switch (T3), and a control unit (14) which coordinates the operation of the controlled supply means and of the activation means in accordance with a predetermined program. The system enables the load to be driven with a particularly low power dissipated.

Description: The present invention relates to a system for driving a reactive load as defined in the preamble of Claim 1 and disclosed in EP-A-0 667 733. Another system for driving a reactive load is disclosed in EP-A-0730392. In a power supply circuit for driving the capacitive load of an electroluminescent (EL) panel the EL panel is connected between the terminals of an autotransformer which has a tap connected to a terminal of a d.c. power supply. A MOSFET is connected between a terminal of the autotransformer and the other terminal of the d.c. power supply. A driving pulse generator circuit is connected to the gate of the MOSFET. The frequency of the driving circuit is made equal to the resonant frequency of the resonant circuit made up of autotransformer and EL panel. As a result, a sinusoidal alternate voltage is supplied to the autotransformer. For simplicity of description, reference will be made below to applications in which the reactive load is capacitive but it is intended that the invention may, in practice, be implemented in just the same manner in applications in which the reactive load is inductive, bearing in mind the equivalence of the voltage and current behaviour of capacitances and inductances. In order to supply energy to a load in a controlled manner, be it a capacitive, an inductive, or a resistive load, it is well known to use an amplifier supplied by a direct-current voltage supply and controlled so as to modulate the supply of a variable quantity of energy to the load in predetermined manner, that is, so as to achieve a given current or voltage waveform in the load. An application of this type with a capacitive load is shown in Figure 1 of the appended drawings. An amplifier 1 has an output stage represented schematically by two controllable current sources G1, G2, connected in series between the rails of a voltage supply, indicated Vs and by the earth symbol. The output terminal of the amplifier, which is the connection node between the two current sources is connected to a capacitive load represented by a capacitor Cl. A control circuit 2 supplies control signals to the amplifier so as to modulate the supply or absorption of current by the current sources G1 and G2, and hence the supply to the load Cl, in accordance with a predetermined program. It is assumed that current is supplied to the load Cl so as to achieve therein a triangular voltage waveform as shown in Figure 2, that is, that the capacitor Cl is to be charged from 0 to a voltage V1, starting from a time t0, in a period t1-t0 and that it is to be discharged in a period t2-t1. The control circuit 2 will therefore activate the current source G1 from the time t0 to the time t1 with the current source G2 deactivated, and will then activate the current source G2 until the time t2 with the current source G1 deactivated. A graph of the current I in the load Cl as a function of time is shown in Figure 3 and a graph of the power Pd dissipated in the current sources G1 and G2 as functions of time is shown in Figure 4. It can easily be shown that, in a practical embodiment, if Cl = 2µF, t1-t0 = 6µs, t2-t1 = 4µs, V1 = 35V and Vs - 40V, there is a constant charge current I1 = 11.6A, a constant discharge current I2 = 17.5A, an instantaneous maximum power P1 = 464W dissipated in the current source G1, an instantaneous maximum power P2 = 612.5W dissipated in the current source G2, a mean power in the period t1-t0 of 156.6 W, a mean power in the period t2-t1 of 122.5 W, and a total power dissipated in the period t2-t0 of 272.1W. In many applications, the power dissipation of the system described above is considered excessive. A need has therefore arisen for alternative systems for driving a reactive load with lower power dissipation. The system according to the invention as defined in general in Claim 1 satisfies this requirement. The invention will be understood better from the following detailed description of some embodiments thereof given by way of non-limiting example, with reference to the appended drawings, in which: Figure 1 shows schematically a known system for driving a capacitive load, Figures 2, 3 and 4 are graphs showing the voltage, the current, and the power dissipated in the system of Figure 1 as functions of time, respectively, Figure 5 is a diagram, partially in block form, of a system for driving a capacitive load according to the invention, Figure 6 is a diagram of a resonant circuit used to illustrate the operation of the system of Figure 5, Figure 7 is a graph of the current in the circuit of Figure 6 as a function of time, Figure 8 shows how a portion of the graph of Figure 7 should be modified to represent the graph of the current produced by the system of Figure 6 according to the invention, Figures 9A-9D are various graphs of voltage, current and power dissipated which are useful for an understanding of the operation of the system according to the invention, Figure 10 shows a second embodiment of the system according to the invention, part of which is similar to that of Figure 5, with some portions shown in greater detail, and Figure 11 shows a third embodiment of the system according to the invention. In the circuit diagram of Figure 5, a capacitive load, for example, a piezoelectric printing head of an ink-jet printer or an element of an electroluminescent panel, again indicated Cl, is connected to the output of an operational amplifier 3 having an output stage represented by two controllable current sources G1 and G2 connected in series with one another between the terminals Vs and earth of a direct-current voltage supply, as in the system of Figure 1. In this case, however, in parallel with the load Cl, there is a circuit constituted by an inductance Lr, an electronic switch T3, for example, a transistor, and a capacitor Cr, connected in series. When the electronic switch T3 is closed, the components Lr and Cr, together with the load Cl, constitute a resonant circuit. An activation unit 10 is connected to the control terminal of the electronic switch T3 in order to open it or close it at predetermined time intervals, as will be explained further below. The operational amplifier 3 has an inverting input connected to a sensor 12 for detecting an electrical quantity in the load and a non-inverting input connected to a digital-analogue convertor or DAC 16. In this embodiment, the sensor is a resistive divider connected in parallel with the load Cl and formed by two resistors R1 and R2. The intermediate tap of the divider is connected to the inverting input of the amplifier 3 in order to supply a voltage thereto, as the electrical quantity indicative of the operation of the resonant circuit. In this embodiment, the DAC 16 also supplies to the amplifier 3 a voltage, more precisely, a voltage which varies as the waveform to be produced in the load Cl. The operational amplifier operates as a comparator of the voltages applied to its inputs and, together with the sensor 12, constitutes a system with error-compensation feedback. The waveform is stored in digital form in a control unit 14 which has the function of coordinating the operation of the system in accordance with a predetermined program. In particular, it is connected to the activation unit 10 in order to provide it with the control signals for the switch T3 at predetermined times correlated with the waveform stored, for example, as a result of the recognition, in the control unit 14, of a sample of the waveform stored which defines a reference moment. In this embodiment, the control unit 14 is formed in a manner such that the waveform stored and the operating program of the system can be modified according to requirements by means of an input unit 15. In other applications, however, it may suffice for the control unit 14 to contain a non-modifiable waveform and a fixed operating program. Reference will be made first of all to Figures 6, 7 and 8 to explain the operation of the circuit according to the invention, shown schematically in Figure 5. Figure 6 shows a resonant circuit constituted by the same components Cr, Cl, Lr and T3 which are present in the circuit of Figure 5. It is assumed that the capacitor Cr is charged to a predetermined voltage and that, at the time t0, the switch T3 which, up to this time has been open, is closed. A sinusoidal current Ir which mirrors the exchange of energy between the inductance Lr and the capacitance of the two capacitive components Cr and Cl flows in the resonant circuit, as shown in Figure 7, and is damped over time because of the internal resistance of the circuit. With reference again to the circuit of Figure 5 according to the invention, it is assumed that, initially, Cr and Cl are discharged and the switch T3 is open. The capacitive load Cl starts to be charged by means of the current source G1 of the operational amplifier 3 until a voltage determined in the control unit 14 is reached and is applied in analogue form to the non-inverting input of the operational amplifier 3 by means of the DAC 16. The control unit 14 then closes the switch T3 by means of the activation unit 10, permitting a transfer of charge from Cl to Cr through the inductance Lr. The transfer of charge finishes when the switch T3 is re-opened at a moment determined by the activation unit 10. According to the invention, the energy stored in the reactive components of the resonant circuit is used, in combination with that supplied or absorbed by the current sources G1 and G2 of the operational amplifier 3, to produce a predetermined waveform in the capacitive load Cl. In this embodiment, in the time interval t0-t1 corresponding to one half period of the sinusoidal current Ir, the waveform is required to be a slope like that of Figure 2 in the same period t0-t1. The first half-wave of the current of Figure 7 is therefore "squared" in order to become identical to that of Figure 3 between t0 and t1. This "squaring" operation can be represented geometrically with reference to Figure 8, if the current in the capacitive load Cl is controlled by means of the operational amplifier 3 in a manner such as to "take away" the top portion of the half-wave, that is, the portion indicated A- in which Ir is greater than I1 and to "add" to the sides of the half-wave, the substantially triangular portions, indicated A+, which are lacking, in order to produce a square wave of amplitude I1. This operation is performed by the system according to the invention shown in Figure 5 under the control of the control unit 14. More particularly, by enabling the activation unit 10, the control unit 14 causes the switch T3 to be closed and applies a reference voltage to the non-inverting input of the operational amplifier 3 by means of the DAC 16. Upon the assumption that the capacitor Cr is already charged, a current due to the operational amplifier 3 and to the energy exchange between the reactive components of the resonant circuit flows in the capacitive load and a corresponding voltage is detected by the sensor 12. The operational amplifier 3 compares this voltage with the reference voltage supplied by the DAC 16. In the embodiment described, after the time t0 at which the switch T3 is closed, as long as the current Ir is less than the constant current I1 required to form the desired voltage slope in the capacitive load Cl, the amplifier 3 supplies the quantity of current which is lacking in order to reach the level I1, by means of the current source G1. As soon as the current in the resonant circuit tends to exceed the value I1 at the time t11, the operational amplifier 3 absorbs the quantity of current in excess of the value I1 by means of the current source G2, discharging it to earth until the time t12. During the subsequent interval from t12 to t1, the operation of the operational amplifier 3 is similar to that in the period from t0 to t11. At the time t1, the activation unit 10 opens the switch T3. Since the time t1 corresponds to the zero-crossing of the current Ir, the activation unit 10 advantageously performs this operation automatically by means of a zero-crossing detector, as shown in Figure 10, which will be described below. In order to evaluate the power dissipated in the driving system according to the invention shown in Figure 5, reference is made to Figure 9B in comparison with Figure 9A which gives the graphs of Figures 2, 3 and 4, restricted to the period t0-t1. The power Pd dissipated is obtained as the integral over time of the product of the voltage and the current. In the known system, power is dissipated throughout the period in which the current source G1 of the amplifier 1 supplies current to the load Cl, that is, from t0 to t1. The amount of power dissipated decreases linearly from a maximum Pd1 when the difference between the supply voltage and the voltage in the capacitive load is at a maximum at the time t0, to a minimum value when G1 ceases to supply current. Since the current I is constant at the level I1, the area AP between the straight line which represents the power dissipated Pd and the coordinate axes is proportional to that AV defined between the straight line which represents the voltage V and the supply-voltage level Vs. In the system according to the invention, as can be seen in Figure 9B, the power dissipated at the time t0 is equal to that dissipated at the time t0 in the known system, but decreases rapidly, since it also benefits from the contribution of the current circulating in the resonant circuit, until it reaches zero at the time t11 when the current Ir in the resonant circuit reaches the level I1 necessary to achieve the desired voltage slope in Cl. In the period between t11 and t12, it is then limited to the contribution of the excess current flowing through the current source G2 and, at the time t12, rises again but to a level lower than its maximum value at the moment t0. It then falls rapidly until it reaches zero at the time t1 at which T3 opens. As can be seen from a comparison between the graphs of the power dissipated in Figures 9A and 9B, the area defined by the curve of the power dissipated Pd and by the coordinate axes is much smaller in Figure 9B than in Figure 9A; this means that the total power dissipated in the period t0-t1 in the system according to the invention is much less than that dissipated in the known system. Upon completion of the charging operation described above, the capacitive load can be discharged in controlled manner, possibly after a waiting period. The discharge of the capacitive load can be controlled by the control unit 14 and by the activation unit 10 by a process similar to the charging process, so as to achieve, in the load, a waveform of opposite sign which may be the same as the charging waveform or different, according to the programming of the control unit 14. The charging and discharging process, with any intervals, can then be continued in accordance with the program of the control unit 14. In order further to reduce the total power dissipated, in an embodiment of the invention shown in Figure 10, in which elements identical to those of Figure 5 are indicated by the same reference numerals, a phase shift is created between the current Ir circulating in the resonant circuit and the charging or discharging current in the capacitive load Cl. This is achieved by delaying or advancing the closure of the switch T3 relative to a predetermined moment within the period of the waveform, according to the waveform to be reproduced in the capacitive load Cl. This phase shift is achieved by means of a suitable delay unit Δt in the control unit 14. In the embodiment described with reference to Figure 9C, the phase shift is an advance, the effect of which can be appreciated from a comparison of Figures 9B and 9C. As can be seen in the voltage graph of Figure 9B, of the voltage contribution to the power dissipated Pd, expressed graphically by the area AV1, in the period in which the current source G1 supplies current before the time t11, is considerable and is greater than that, expressed by the area AV2, in the period following the time t12 in which the current source G1 supplies current again. A small advance Δt in the closure of T3, as shown in Figure 9C, reduces the both the voltage and the current contributions in the period t0-t11 so that the net result is a reduction in the mean power dissipated. A further reduction in the mean power dissipated is achieved by reducing the supply voltage of the operational amplifier 3 during the period of time in which the voltage in the capacitive load is low, for example, by changing from a level Vs to a level Vs/2, as shown in Figure 9D. As can be seen in the voltage graph, the contribution of the voltage to the determination of the power'in the period of the initial charging of Cl is further reduced in this case. This effect is achieved by means of a suitable supply with two switchable voltage levels. A schematic example of a supply of this type is shown in Figure 10. A voltage supply 38 with two output levels is connected to the supply terminals of the operational amplifier 3 in parallel with a smoothing capacitor C. An electronic switch T4 is associated with the supply 38 and is controlled by the output of a comparator 37. The latter has one input connected to a reference supply VREF and the other input connected to the sensor 12. When the voltage detected by the sensor is equal to or less than VREF, the output of the comparator is at a low level, that is, such as not to activate the switch T4 and the operational amplifier 3 is supplied by the lower-level voltage. When the level of the voltage detected is greater than VREF, however, the output of the comparator 37 is at a high level such as to close the switch T4 so that the operational amplifier 3 is supplied with the higher-level voltage. Since, as can be seen in the voltage graph of Figure 9D, the contribution of the voltage to the power dissipated is reduced in the period in which the current source G1 supplies current for the initial charging of the load Cl, the power dissipated in this period of time is also reduced. Naturally, the two measures described above for further reducing the power dissipated may also be used individually. Figure 10 also shows in some detail the unit 10 for activating the switch T3. This unit 10 comprises a flip-flop 30 an input S of which receives a switching (setting) signal from a counter 33 connected to the control unit 14. The time at which this signal is emitted is determined by the control unit 14 on the basis of the wave-form programmed to be reproduced in the capacitive load and on the basis of any delay or advance Δt programmed. The setting signal at the input S of the flip-flop produces a "high" signal at the output Q of the flip-flop such as to close the switch T3. The counting cycle of the counter is selected so as to define the most suitable moment to close the switch T3 within the period of the waveform. The activation unit 10 also comprises a current zero-crossing detector 32 comprising a capacitor Co and a resistor Ro connected in series with one another in parallel with the capacitor Cr of the resonant circuit, as well as two diodes connected so as to conduct in opposite directions in parallel with the resistor Ro, in order to limit the voltage drop therein. When the switch T3 is closed, a sinusoidal voltage which is in phase with the current passing through the inductance Lr appears in the resistor Ro. Every zero crossing of the current passing through the resistor Ro is detected by a zero-crossing detector and indicator circuit, indicated by a block 32. The circuit 32 emits an output pulse which zeroes the counter 33 and causes the flip-flop 30 to switch to the "reset" state R, thus causing the switch T3 to open. In Figure 10, the inductance Lr is represented by three separate windings, of which one is in counterphase, and which are connected as shown in order to absorb and discharge the recirculating current which is created during the opening and closure of the switch T3. In a practical embodiment of the system according to the invention for driving a piezoelectric ink-jet printing head, in which the capacitive load was equal to that of the known application described at the beginning with the same voltages and times (Cl = 2µF, t1 = 6µs, V1 = 35V and Vs = 40V) and with an inductance Lr = 4.5µH, a capacitor Cr = 1.4µF and a delay Δt = 0.8µs, a mean power dissipated of 20W was obtained, that is, much less than that obtained with the known driving system. The embodiment of the invention shown in Figure 11, in which components identical to those of Figure 10 are indicated by the same reference symbols or numerals, relates to an application in which the capacitive load to be driven may vary, that is, may adopt different capacitances of predetermined values. Various inductances are provided and can be connected in the circuit selectively in order to optimize the energy balance between the various reactive components of the resonant circuit in any situation; in this embodiment, three inductances are provided but, naturally, there may be a larger number, if necessary. The three inductances, indicated Lr1, Lr2 and Lr3, are connected in the resonant circuit in series with three respective electronic switches, indicated T31, T32, T33, each having its control terminal connected to the output of a respective flip-flop 301, 302 or 303 of a control unit 10' similar to the unit 10 of Figure 10. The capacitance value of the capacitive load at any particular time, indicated Clv in this example, is determined by the input unit 15 and is supplied to the control unit 14. By means of the activation unit 10', the control unit 14 selects one of the electronic switches T31, T32, T33, more precisely, the switch which is in series with the appropriate inductance for the capacitance value Clv of the load, as well as the magnitude of the delay Δt most suitable for the specific combination of capacitance and inductance and for the waveform to be produced in the capacitive load. Naturally, it is also possible to provide for the possibility of excluding the activation of the resonant circuit by keeping all of the electronic switches permanently open. In this case, the system functions exactly like the known system. The selection of this type of operation may be appropriate for capacitive loads of very low value, that is, when the advantages of the resonant circuit would not be appreciable. Although only a few embodiments of the invention have been described and illustrated, it is clear that many variants are possible within the scope of the invention as defined by the appended claim. For example, as already mentioned at the beginning, it would be possible to control an inductive load rather than a capacitive load; in this case, it would be necessary to detect the voltage zero-crossing instead of the current zero-crossing. Moreover, a parallel resonant circuit could be used instead of a series resonant circuit. Furthermore, it would clearly be possible to produce any current waveform stored in the control unit 14, rather than a constant-current waveform.

Claims: A system for driving a reactive load (Cl), comprising: means (3, G1, G2, 12) for the controlled supply of the load (Cl), for supplying variable quantities of energy to the load (Cl) in a predetermined manner, at least one reactive component (Cr, Lr), a controllable electronic switch (T3), means (10) for activating the electronic switch (T3), and a control unit (14) connected to the controlled supply means (3, G1, G2, 12) and to the activation means (10) in order to coordinate their operation in accordance with a predetermined program which defines the supply of energy to the load, characterized in that said at least one reactive component (Cr, Lr) is connected to the load (Cl) by means of the controllable electronic switch (T3) and forms a resonant circuit with the load when the electronic switch is closed, the control unit (14) generates a reference quantity corresponding to the energy to be supplied to the load, and the controlled supply means (3, G1, G2, 12) comprise a sensor (12) for detecting an electrical quantity of the load and error-compensation means (3, G1 G2) which can control the electrical quantity detected so as to keep it equal to the reference quantity. A system according to Claim 1, comprising input means (15) associated with the control unit (14) for modifying the program which defines the supply of energy to the load. A system according to Claim 1 or 2, in which the error-compensation means (3, G1, G2) comprise a comparator (3) which receives the electrical quantity detected and the reference quantity as inputs in order to compare them, and an output of which is connected to the load (Cl) in order to supply energy or to absorb energy in dependence on the outcome of the comparison. A system according to Claim 3, in which the comparator comprises an operational amplifier (3). A system according to any one of the preceding claims, in which the means (10) for activating the electronic switch (T3) comprise a detector (32) for detecting the zero-crossing of a sinusoidal electrical quantity in phase with an electrical quantity in the resonant circuit and means (30) for opening the electronic switch (T3) when the detector detects a zero-crossing of the sinusoidal electrical quantity. A system according to any one of the preceding claims, in which the reactive load is a capacitive load (Cl) and the resonant circuit formed by the capacitive load and by at least one reactive component comprises an inductance (Lr) and a capacitance (Cr) in series with one another. A system according to any one of the preceding claims, in which the controlled supply means (3, G1, G2, 12) comprise a multi-level voltage supply, and in which means (37, T4, 12) are provided for selecting the supply-voltage level in dependence on an electrical quantity in the resonant circuit. A system according to Claim 7, in which the means for selecting the supply-voltage level comprise the sensor (12) for detecting an electrical quantity.

Inventors: PIDUTTI ALBINO, SCURATI MARIO, PIDUTTI, ALBINO, SCURATI, MARIO
Applicants: ST MICROELECTRONICS SRL, STMICROELECTRONICS S.R.L.
Application year: 1997
Main classification: H02J1
IPC codes: B41J2, H05B33, B41J29