Charge pump for a semiconductor substrate

Abstract: A low voltage current source generates low voltage signals for powering a variable frequency oscillator. The low voltage signals are at a slightly higher voltage until a negative substrate bias is achieved. The oscillator operates at a low frequency for low power consumption when no charge pumping is needed and at a higher frequency when charge pumping is in fact needed or when charge pumping will most likely be needed. The variable frequency oscillator controls a timing signal generator which generates the timing signals used to control the overall operation of the charge pump system. Voltage translation circuitry translates the low voltage current source signals into higher voltage signals which are used to translate the substrate voltage from its negative value to a positive value so that the substrate voltage may be compared to a reference voltage using a conventional comparator. When the substrate voltage is above the desired level, the comparator generates a pump activating signals to a pump signal generator which, in turn, generates the necessary signal to cause the charge pump to operate.

Description: This invention relates to an apparatus for translating voltages according to claim 1. MOS transistors are commonly used in electronic circuits such as dynamic random access memories (DRAMS). In an NMOS transistor, an N-type source region is separated from an N-type drain region by a P-type channel region. All three regions are formed in a P-type semiconductor substrate. By applying a positive voltage to a gate electrode disposed above the channel region, electrons gather in the channel region between the source region and the drain region to allow current to flow from the drain region to the source region. PMOS transistors have the same structure except the conductivity types of the various regions are reversed and a negative gate voltage is required to allow current to flow from the source region to the drain region. It has been found that NMOS transistors operate better when the P-type substrate of the NMOS (or of the NMOS transistors in a CMOS circuit) is driven negative with respect to circuit ground, in other words there is a negative substrate bias. Such a negative substrate bias provides a number of advantages in terms of the overall circuit performance. More specifically, a negative substrate bias decreases the NMOS transistor source and drain capacitance, decreases the likelihood of latchup, decreases PN diode injection when a node is driven below ground, and decreases the effective body effect, all of which are desirable in CMOS circuits. Typically a charge pump circuit is used to create the negative substrate bias. Once a negative substrate bias is achieved, however, it does not last forever. For example, when an NMOS transistor is conductive with a relatively high drain to source voltage, some of the electrons traveling from the source region to the drain region collide with atoms in the channel region with enough energy to cause electron/hole pairs to form. The positive gate voltage attracts the generated electrons to the surface of the channel while the positive drain voltage attracts them to the drain where they simply add to the normal flow of electrons from source to drain. The positively charged holes, by contrast, are repelled by the positively charged gate away from the channel region into the substrate. The substrate current created by the excess holes makes the substrate more positively charged, thus counteracting the negative substrate bias. In DRAMS, a substantial amount of substrate current is generated whenever the memory is read or written, since many transistors are switched on and off at that time. This component of substrate current may be orders of magnitude above the background (i.e., standby) leakage current of all the reverse biased P-N diodes throughout the circuit. Therefore, the charge pump must remove low substrate current during standby and high substrate current during high activity to maintain the negative substrate bias. Figure 1 is a conceptual schematic diagram of a charge pump 2 which includes a first switch 4 coupled between a positive power supply voltage (V cc ) and a first terminal 6 of a capacitance C1. A second switch 8 is coupled between a ground potential (V ss ) and a second terminal 10 of capacitance C1. A third switch 12 is coupled between (V ss ) and terminal 6 of capacitance C1, and a fourth switch 14 is coupled between the substrate (represented by the voltage (V bb )) and terminal 10 of capacitance C1. In operation, switches 4 and 8 are both closed (made conductive) for charging capacitance C1 to a voltage equal to the difference between (V cc ) and (V ss ). In Figure 1, (V cc ) = +5 volts and (V ss ) = 0 volts, so capacitance C1 charges with node 6 five volts more positive than node 10. Thereafter, switches 4 and 8 are opened and switches 12 and 14 are both closed. Since the positive terminal 6 of capacitance C1 is now coupled to a ground potential, the negative terminal 10 of capacitance C1 tries to drive V bb to negative 5 volts through switch 14. Thereafter, switches 12 and 14 are opened, and the sequence repeats itself. An oscillator (not shown) typically controls the repetitive switching sequence, and a detector (not shown) monitors the substrate voltage and controls the pumping operation to maintain the substrate at the proper negative voltage level. As discussed in more detail below, known charge pumps consume a substantial amount of power (often 1 milliwatt or more even when no further pumping is required), often work against themselves by adding positive substrate current as they operate, and generally operate inefficiently. Voltage boosting circuitry is known in the art. For instance, a pair of input signals can be used to generate a substantially constant output voltage that is higher than the maximum voltage of either input signal, as described in EP-A-0 576 008. The present invention is directed to voltage translation circuitry capable of translating a negative voltage into a positive voltage signal (e.g., between 0 and +5 volts), without adding substrate current as it operates. This circuitry allows a negative voltage to be compared to a positive reference voltage using a conventional comparator. The invention can be advantageously employed in reducing the power consumption of a charge pump used to maintain a negative substrate voltage. After the voltage translation circuitry translates the negative substrate voltage to a positive value, a comparator compares this translated voltage to a positive reference voltage that corresponds to the desired level of the substrate voltage. When the substrate voltage is more positive than the desired level, the comparator transmits a pump activating signal to a pump signal generator which turns on the charge pump. When the substrate voltage is more negative than the desired level, a "slow pump" signal is transmitted instead, and the charge pump may be turned off to reduce power consumption. A better understanding of the nature and advantages of the charge pump circuit of the present invention may be had with reference to the detailed description and the drawings below. Figure 1 is a conceptual schematic diagram of a known charge pump; Figure 2 is a block diagram of a particular embodiment of a charge pumping system according to the present invention; Figure 3 is a schematic diagram of a timing signal generator shown in Figure 2; Figure 4 is a timing diagram illustrating the timing of signals generated by the timing signal generator shown in Figure 9; Figure 5 is a schematic diagram of a known substrate voltage comparator; Figure 6 is a conceptual schematic diagram of a particular embodiment of a substrate voltage detector according to the present invention; Figure 7 is a schematic diagram of a particular embodiment of the logic voltage level translator shown in Figure 2; Figure 8 is a schematic diagram of particular embodiments of the substrate voltage translator and comparator shown in Figure 2; Figure 2 is a block diagram of a particular embodiment of a substrate charge pumping system 20 according to the present invention. A low voltage generator 24 provides a reduced potential source (for power saving) through bus 32 to power a variable (e.g., dual) frequency oscillator 28 and timing signal generator 34. Variable frequency oscillator 28 provides oscillation signals of high or low frequency through a bus 36 to timing signal generator 34. High frequency is for fast pumping and low frequency is for reduced power. In response, timing signal generator 34 provides timing signals to control the operation of the remaining portions of the circuit. In particular, generator 34 provides timing signals to logic voltage level translator 40, comparator 56 and substrate voltage translator 44 through bus 48. Logic level voltage translator 40 translates the low voltage signals, for example 0 to +1.5 volts, produced by oscillator 28 and timing generator 34 into higher voltage signals, of for example 0 to +5 volts. The higher voltage signals are supplied to comparator 56 through a bus 50. Substrate voltage translator 44 translates the substrate voltage from a level below ground to a level above ground and supplies the translated voltage to comparator 56 through a bus 58. Comparator 56 compares the translated voltage received on bus 58 to a reference voltage received on a bus 60 and provides, when needed, pump activating signals to a pump signal generator 64 through a bus 68. Comparator 56 also provides complimentary SPUMP/NSPUMP ( S low pump/ N ot S low pump = Low Frequency/Not Low Frequency) signals on SPUMP signal line 70 and NSPUMP signal line 72 to oscillator 28. The SPUMP and NSPUMP signals are used to control the frequency of the oscillator. Pump signal generator 64 generates timing signals on a bus 76 for controlling the operation of a charge pump 80. Charge pump 80 creates and maintains a negative substrate bias. Charge pump 80 uses special circuitry which does not experience an undesirable threshold drop in any diode-connected transistor. Figure 3 is a schematic diagram showing the construction of timing signal generator 34 and how it is coupled to dual frequency oscillator 28. To save power, all of the circuitry shown in Figure 3 operates preferably from the reduced power supply voltage provided by low voltage generator 24. Timing signal generator 34 includes an inverter 300 having an input terminal coupled to the output terminal of inverter 84A in oscillator 28 and an output terminal coupled to an input terminal of an inverter 304. The output terminal of inverter 304 is coupled to one input terminal of a 2-input NAND gate 308. The other input terminal of NAND gate 308 is coupled to the output terminal of an inverter 312 which, in turn, has an input terminal coupled to an output terminal of inverter 84C in oscillator 28. Similarly, an inverter 316 has an input terminal coupled to the output terminal of inverter 84B in oscillator 28 and an output terminal coupled to an input terminal of an inverter 320. The output terminal of inverter 320 is coupled to one input terminal of a 2-input NAND gate 324. The other input terminal of NAND gate 324 is coupled to an output terminal of an inverter 328 which, in turn, has an input terminal coupled to an output terminal of inverter 84D in oscillator 28. The output terminal of NAND gate 308 is coupled to an input terminal of an inverter 332, to one input terminal of a 2-input NAND gate 310, and to one input terminal of a 2-input NAND gate 356. The output terminal of inverter 332 is coupled to an input terminal of an inverter 336 which has an output terminal connected to an input terminal of an inverter 340. An output terminal of inverter 340 is coupled to an input terminal of an inverter 344 which has an output terminal coupled to an input terminal of an inverter 348. An output terminal of inverter 348 is coupled to an input terminal of an inverter 352 which has an output terminal coupled to the other input terminal of NAND gate 356 and to an "X" signal line 354. An output terminal of NAND gate 356 is coupled to a input terminal of an inverter 360 which has an output terminal coupled to an input terminal of an inverter 364. An output terminal of inverter 364 is coupled to a "Y" signal line 368. An output terminal of NAND gate 310 is coupled to one input terminal of a 2-input NAND gate 370 and to an input terminal of an inverter 374. The other input terminal of NAND gate 370 is coupled to an output terminal of NAND gate 324. An output terminal of inverter 374 is coupled to an input terminal of an inverter 378, and an output terminal of inverter 378 is coupled to a "Z" signal line 382. An output terminal of NAND gate 370 is coupled to the other input terminal of NAND gate 310 and to an input terminal of an inverter 390. An output terminal of inverter 390 is coupled to an input terminal of an inverter 394, and an output terminal of inverter 394 is coupled to a "W" signal line 398. "X" signal line 354, "Y" signal line 368, "Z" signal line 382, and "W" signal line 398 together comprise bus 48 (Figure 2). Figure 10 is a timing diagram showing the sequence of signals on "X" signal line 364, "Y" signal line 368, "Z" signal line 382, and "W" signal line 398, respectively. In general, the signals on the "X" signal line 354 and on the "Y" signal line 368 are complimentary, but with timing such that the signal on "X" signal line 354 goes high before the signal on "Y" signal line 368 goes low and vice versa. The same is true for the signals on "Z" signal line 382 and "W" signal line 398. That is, the signals on the lines each have a high portion (+1.5 volts) and a low portion (0 volts), and the low portions of the signals are mutually exclusive. Figure 7 is a schematic diagram of a particular embodiment of logic voltage level translator 40. "X" signal line 354, Y signal line 368, "W" signal line 398 and "Z" signal line 382 are received from timing signal generator 34. The logic voltage level translator receives logic input signals "X", "Y", "W" and "Z", in which the low logic level is V ss (0 volts) and the high logic level is about 1.5 volts. Its purpose is to provide output signals which switch between V ss and V cc . NMOS transistor 450, functioning as a capacitor, has both its source and drain terminals connected to "X" signal line 354 and its gate terminal 461 connected to a node 488. NMOS transistor 452, also functioning as a capacitor, has both its source and drain terminals connected to "Y" signal line 368 and its gate terminal 463 connected to a node 440. NMOS transistor 454, also functioning as a capacitor, has both its source and drain terminals connected to "W" signal line 398 and its gate terminal connected to a node 650. NMOS transistor 456, also functioning as a capacitor, has both its source and drain terminals connected to "Z" signal line 382 and its gate terminal 560 connected to a node 652. A PMOS transistor 460 has a source terminal 464 coupled to (V cc ), a drain terminal 468 coupled to another terminal 461 of capacitance 450 and a gate terminal 472 coupled to node 490. A PMOS transistor 476 has a source terminal 480 coupled to (V cc ), a drain terminal 484 coupled to node 490 and a gate terminal 482 coupled to node 488. An NMOS transistor 500 has a drain terminal 504 coupled to (V cc ), a source terminal 508 coupled to node 488 and a gate terminal 512 coupled to a (V cc -|V tp |) volt signal. Similarly, an NMOS transistor 520 has a drain terminal 524 coupled to (V cc ), a source terminal 528 coupled to node 490, and a gate terminal 532 coupled to the (V cc -|V tp |) volt signal. . A PMOS transistor 550 has a source terminal 554 coupled to (V cc ), a gate terminal 558 coupled to node 652, and a drain terminal 564 coupled to node 650. A PMOS transistor 572 has a source terminal 574 coupled to (V cc ), a drain terminal 576 coupled to node 652, and a gate terminal 580 coupled to node 650. An NMOS transistor 600 has a drain terminal 604 coupled to (V cc ), a source terminal 608 coupled to node 650, and a gate terminal 612 coupled to the (V cc -|V tp |) volt signal. An NMOS transistor 630 has a drain terminal 634 coupled to (V cc ), a source terminal 638 coupled to node 652, and a gate terminal 642 coupled to the (V cc -|V tp |) volt signal. A PMOS transistor 660 has a source terminal 664 coupled to (V cc ), a drain terminal 668 coupled to a node 672, and a gate terminal 676 coupled to node 488. An NMOS transistor 680 has a drain terminal 684 coupled to node 672, a source terminal 688 coupled to (V ss ), and a gate terminal 692 coupled to "X" signal line 354. Node 672 is coupled to an input terminal of an inverter 700 having an output terminal coupled to an input terminal of an inverter 704. An output terminal of inverter 704 is coupled to a line 706 which provides signals to substrate voltage translator 44. Similarly, a PMOS transistor 710 has a source terminal 714 coupled to (V cc ), a drain terminal 718 coupled to a node 720, and a gate terminal 724 coupled to node 490. An NMOS transistor 730 has a drain terminal 734 coupled to node 720, a source terminal 738 coupled to (V ss ), and a gate terminal 742 coupled to "Y" signal line 368. Node 720 is coupled to an input terminal of an inverter 750 which has an output terminal coupled to an input terminal of an inverter 754. An output terminal of inverter 754 is coupled to a line 760 which provides signals to substrate voltage translator 44. Lines 650, 706, and 760 together comprise bus 50 (Figure 2). The function of the part of the logic voltage level translator 40 shown in the left hand part of Figure 7 is to generate signals that transition from V ss to V cc on nodes 706 and 760 from the low voltage signals "X" and "Y" on signal lines 354 and 368, respectively. The function of the remainder of the logic voltage level translator 40 is to generate a signal on node 650 that transitions between V cc -1.5 volts and V cc from the low voltage signals "W" and "Z". Both the left and right portions of the circuitry of Fig.7 work in the same manner. Nodes "X" on signal line 354 and "Y" on signal line 368 transition between V ss (0 volts) and 1.5 volts, and are generally complements of one another. That is, when one is low the other is high and visa versa. But timing signal generator 34 of Figure 2 shown in detail in Figure 3 provides that node "X" will transition high before mode "Y" transitions low and that node "Y" will transition high before node "X" transitions low, as seen in Figure 4. Referring again to Figure 7, when node "Y" transitions low, capacitor (NMOS transistor) 452 drives node 490 low turning on PMOS transistor 460, and pulling node 488 to V cc . Throughout this time, node "X" is high at +1.5 volts, and capacitor 450 is charged to 3.5 volts. Next, node "Y" transitions high to +1.5 volts, capacitor 452 drives node 490 high, turning off PMOS transistor 460, but with node 488 remaining at V cc . Next, node "X" transitions low from 1.5 volts to 0 volts and capacitor 450 drives node 488 down by 1.5 volts to 1.5 volts below V cc , turning on PMOS transistor 476. This pulls node 490 up to V cc at a time when node "Y" is high at +1.5 volts, charging capacitor 452 to 3.5 volts. When node "X" then transitions high, node 488 is again returned to V cc , turning off PMOS transistor 476, but with node 490 remaining at V cc . This completes one full cycle. Thus a high level of V cc exists on node 488 when node "X" is high at +1.5 volts and a low level of V cc -1.5 volts exists on node 488 when node "X" is at a low level of 0 volts. When node "X" is high at +1.5 volts, NMOS transistor 680 is on and pulls the input of inverter 700 to 0 volts. At this time, node 488 is at V cc so PMOS transistor 660 is off and no current flows through transistors 660 and 680. When node "X" is low at 0 volts, node 488 is low at V cc -1.5 volts, turning on PMOS transistor 660 and pulling the input of inverter 700 to V cc . At this time, NMOS transistor 680 is off and again no current flows through transistors 660 and 680. Thus, this circuitry generates on the input of inverter 700 a full logic swing between V ss and V cc from low level inputs on nodes "X" and "Y", and does so without establishing any current path from V cc to V ss . Logically, output 706 of the logic voltage level translator of Figure 13 is a high level compliment of the low level signal on node "X". If node "X" instead went directly to a normal inverter whose PMOS source voltage was +5 volts, the inverter would consume substantial power whenever node "X" is at +1.5 volts. In an identical manner, the low-level signal on node "Y" generates its compliment as a high level signal on node 760. NMOS transistors 500 and 520 are used to start the circuit when power is first applied, and are not needed or functional thereafter. Each establishes a voltage of at least V cc -|V tp |-Vtn = V cc -1./5 volts on its respective node 488 or 490, sufficient to establish a channel in each of NMOS transistors (capacitors) 450 and 452, which in turn causes the circuit to function as described above. The circuitry on the right side of Fig. 7 translates the 0 to +1.5 volt logic levels on node "W" to V cc -1.5 to V cc volt logic levels on node 650. These levels on node 650 will be used to turn on or off a PMOS transistor whose source is at V cc . This circuitry functions in an identical manner to the corresponding portions of the circuitry on the left side of Figure 7 already described. A discussion of a known substrate voltage detector is in order to appreciate the inventive aspects of some of the remaining portions of the charge pumping system of the present invention. Figure 5 is a schematic diagram of a known substrate voltage comparator 300. Voltage comparator 300 includes a very narrow, very long channel PMOS transistor 304 having a source terminal 308 coupled to (V cc ), a gate terminal 312 coupled to (V ss ), and a drain terminal 316 coupled to a node 320. Transistor 304 functions as a very high resistance or very low current source. An NMOS transistor 324 has a drain terminal 328 coupled to node 320, a gate terminal 332 coupled to a reference voltage V ref (typically ground), and a source terminal 336 coupled to a node 340. An NMOS transistor 344 has a drain terminal 348 and a gate terminal 352 together coupled to node 340, and a source terminal 356 coupled to a node 360. Finally, an NMOS transistor 364 has a drain terminal 368 and a gate terminal 372 together coupled to node 360, and a source terminal 376 coupled to the substrate V bb . All the NMOS transistors, 324, 344, and 364 are relatively wide, short channel transistors. If the substrate voltage is more negative than three NMOS threshold voltages below V ref (ground), there will be conduction through transistors 304, 324, 344 and 364 between the positive supply V cc and the negative substrate V bb . In this case, transistor 364 will prevent node 360 from rising above V bb plus one NMOS threshold voltage, and transistor 344 will prevent node 340 from rising more than one NMOS threshold voltage above that, i.e. V bb + 2 (V tn ). With the source voltage of NMOS transistor 324 (V bb + 2(V tn )) its gate voltage V ref is one and transistor 324 pulls node 320 down to essentially the voltage on node 340, i.e. V bb + 2 (V tn ). This voltage, below ground, is one logic state for the output on node 320 signifying that no pumping is necessary. If, on the other hand, V bb is less than (more positive voltage than) 3 NMOS threshold voltages below V ref (ground), then diode connected transistors 364 and 344 cannot pull node 340 low enough to cause conduction through transistor 324. In this case, there is no current through the circuit and PMOS transistor 304 pulls node 320 up to V cc . This V cc level on node 320 is the other logic state for the output on node 320, signifying that pumping is necessary. The signal at node 320 is communicated to the charge pump. The charge pump turns on for transferring charge to the substrate when node 320 is at (V cc ) volts, and the charge pump is turned off when node 320 is at V bb + 2(V tn ). Unfortunately, whenever V bb is low enough to turn off the charge pump, there is current flowing through comparator 300 into the substrate. Thus, the comparator 300 itself causes a substrate current which must be pumped away. The substrate voltage comparison circuitry according to the present invention avoids this problem. Figure 6 is a conceptual schematic diagram of a particular embodiment of substrate voltage comparison circuitry according to the present invention. A switch 400 is coupled between ground (V ss ) and a terminal 404 of a capacitance C2. A switch 408 is coupled between the substrate voltage V bb and a terminal 412 of capacitance C2. A switch 414 is coupled between the power supply voltage (V cc ) and terminal 404 of capacitance C2, and one input terminal 418 of comparator 56 is coupled to terminal 412 of capacitance C2 via bus 58. As noted previously, the other input terminal of comparator 56 is coupled to reference voltage (V ref ) through bus 60. The purpose of the switch and capacitance circuitry shown in Figure 6 is to translate the substrate voltage V bb to a level which may be compared by comparator 56. Initially, switches 400 and 408 are closed for charging capacitance C2 to a voltage equal to (V ss - |V bb |). Thereafter, switches 400 and 408 are opened. Switch 414 is then closed which causes terminal 404 of capacitance C2 to rise from V ss to V cc . This, in turn, causes the voltage at terminal 412 to rise from V bb to V bb + V cc , assuming V ss equals 0 volts. If V cc equals +5 volts, and if V bb is more positive than -5 volts, then the voltage on bus 58 is now a positive voltage which may be conveniently compared by comparator 56. After the voltage on bus 58 is compared by comparator 56, switch 414 is opened and switch 400 is closed. Terminal 404 of capacitance C2 falls to V ss , and terminal 412 falls to V bb . Switch 408 then may be closed with no transfer of charge to or from the substrate. The circuit thus operates without the disadvantage discussed for known comparators. Figure 8 is a schematic diagram of a particular embodiment of substrate voltage translator 44 and comparator 56. Voltage level translator 44 corresponds to the switch and capacitor of Figure 6, while comparator 56 of Figure 8 more or less corresponds to comparator 56 of Figure 6. Referring to the substrate voltage translator 44 portion of Figure 8, line 760 is coupled to source and drain terminals of PMOS transistors functioning as capacitances 804 and 812. Similarly, line 706 is coupled to source and drain terminals of a PMOS transistor functioning as a capacitance 820. An NMOS transistor 830 has a drain terminal 834 coupled to a gate terminal 838 of capacitance 812, a source terminal 842 coupled to the substrate V bb , and a gate terminal 846 coupled to a gate terminal 850 of capacitance 820. An NMOS transistor 854 has a drain terminal 858 coupled to gate terminal 850 of capacitance 820, a source terminal 862 coupled to V bb , and a gate terminal 866 coupled to gate terminal 838 of capacitance 812. The circuit described thus far operates in the same manner as in the circuit shown in Figure 7 except all polarities are reversed. The voltage on gate terminals 838 and 850 swing between V bb and V bb + V cc volts as nodes 760 and 706 swing between V ss (0 volts) and V cc . That is, when terminal 838 is high as a result of a high (e.g., +5 volts) signal on line 760, transistor 854 turns on pulling gate terminal 850 to V bb during which time the signal on line 706 is low. Thereafter, the signal on line 760 goes low turning off transistor 854. Then when the signal on line 706 goes high (e.g., +5 volts), the voltage on gate terminal 850 rises to V bb + 5 volts, turning on transistor 830 and pulling gate terminal 838 to V bb . An NMOS transistor 880 has a drain terminal 884 coupled to a gate terminal 888 of capacitance 804 and to a node 885, a source terminal 892 coupled to a node 896, and a gate terminal 900 coupled to gate terminal 850 of capacitance 820. Another NMOS transistor 904 has a drain terminal 908 coupled to node 896, a source terminal 912 coupled to V bb , and a gate terminal 916 coupled to gate terminal 850 of capacitance 820. Finally, an NMOS transistor 930 has a drain terminal 934 coupled to V cc , a source terminal 938 coupled to node 896, and a gate terminal 942 coupled to node 885. The switches and capacitor of Figure 6 correspond to the following transistors of Figures 7 and 8. Switches 400 and 414 of Figure 6 correspond to the NMOS and PMOS transistors respectively of inverter 754 of Figure 7 which generates node 760. Capacitor CZ of Figure 6 corresponds to capacitor (PMOS transistor) 804 of Figure 8. Switch 408 of Figure 6 corresponds to the series combination of NMOS transistors 880 and 904 of the substrate voltage translator 44 in Figure 8. The translated substrate voltage on bus 58 of Figure 6 corresponds to the translated substrate voltage bus 58 of Figure 8. As previously discussed, the voltages on the gate terminals of capacitors (PMOS transistors) 812 and 820 alternate between V bb and (V bb + V cc ). When node 706 is high, node 760 is low. At this time gate terminal 850 of capacitor 850 is at (V bb + V cc ), turning on transistors 880 and 904, pulling node 885 to V bb . Then when node 706 goes low, the gate terminal 850 of capacitor 820 returns to V bb , and transistors 880 and 904 hopefully turn off. Finally node 760 rises to V cc and node 885 is driven to (V bb + V cc ) as the translated substrate voltage on bus 58 to comparator 56. To maintain integrity of this (V bb + V cc ) voltage between the time node 885 rises and the time comparator 56 is sensed, it is important that the voltage on node 885 not be allowed to leak off. But NMOS transistor 904 has its source at V bb and has no back bias or body effect to raise its threshold voltage as do NMOS transistors whose source is at V ss , well above the V bb potential. Without any body effect, transistor 904 may not shut off completely. That is, it may have a small leakage current even when its gate terminal is at V bb . Transistors 880 and 930 are included to prevent this problem from leaking charge off node 885. When node 885 rises to (V bb + V cc ), source follower NMOS transistor 930 pulls node 896 up to a voltage of [(V bb + V cc ) - V tn ]. Transistor 880, with its source 892 well above V bb and its gate 900 at V bb , is fully off and has totally negligible leakage current. Therefore as node 885 rises to (V bb + V cc ), its level remains intact, and substrate voltage translator 44 provides a voltage level of (V bb + V cc ) into comparator 56. The reference voltage V ref on bus 60 into comparator 56 of Figure 8 may be generated by a simple capacitive divider (not shown). If, for example, two capacitors are in series between ground and a node that switches from ground to V cc , the intermediate node (between the capacitors) will switch by a fraction of V cc depending on the ratio of the capacitances. The intermediate node is discharged to ground (by an NMOS transistor) when the switching node is at ground. Thus the voltage on the intermediate node switches up to a fraction of V cc depending on the capacitance ratio. This reference voltage is compared to the (V bb + V cc ) voltage provided by substrate voltage translator 44. If these capacitors are of equal value, then this intermediate node voltage is V cc /2. In this case the regulator will turn on the charge pump whenever V bb is more positive than -(V cc )/2 and turn it off otherwise. That is, the regulator will strive to maintain a substrate voltage such that the voltage on bus 58 = (V bb +V cc ) = 1/2 (V cc ) = V ref ) or (V bb ) = -1/2 (V cc ). The translated substrate voltage signal on bus 58 is communicated to comparator 56 where it is compared to the reference voltage V ref received on bus 60. The comparison is triggered by the signals on "W" signal line 398 from timing signal generator 34. In this embodiment, the W signals occur once per oscillator cycle to provide the once-per-cycle comparison discussed above. If the substrate voltage is more positive than the reference voltage, then a pump activating signal in the form of a positive pulse appears on line 68. Additionally, the differential SPUMP/NSPUMP signals are generated on SPUMP signal line 70 and NSPUMP signal line 72 upon each comparison and remain valid until the next cycle. The detailed capacity of comparator 56 is shown in Figure 8. The "W" signals on "W" signal line 398 are communicated to a gate terminal 1000 of an NMOS transistor 1004 for executing the comparison function. NMOS transistor 1004 has a source terminal 1008 coupled to V ss and a drain terminal 1012 coupled to a node 1016. Node 1016 is, in turn, coupled to a source terminal 1020 of an NMOS transistor 1024 and to a source terminal 1028 of an NMOS transistor 1032. A gate terminal 1036 of NMOS transistor 1024 is coupled for receiving the reference voltage V ref (which may be generated via a capacitive divider coupled between V cc and V ss as previously discussed) on bus 60, and a gate terminal 1040 of NMOS transistor 1032 is coupled for receiving the translated substrate voltage on bus 58. NMOS transistor 1024 has a drain terminal 1044 coupled to a source terminal 1048 of an NMOS transistor 1052. NMOS transistor 1052 has a gate terminal 1056 coupled to a node 1060 and a drain terminal 1064 coupled to a node 1068. Node 1068 is coupled to a drain terminal 1072 of a PMOS transistor 1076 and to a drain terminal 1080 of a PMOS transistor 1084. A gate terminal 1088 of PMOS transistor 1076 is coupled to line 650, and a gate terminal 1092 of transistor 1084 is coupled to node 1060. A source terminal 1096 of transistor 1076 and a source terminal 1100 of transistor 1084 are both coupled to V cc . A drain terminal 1104 of transistor 1032 is coupled to a source terminal 1108 of an NMOS transistor 1112. NMOS transistor 1112 has a gate terminal 1116 coupled to a node 1120 (which is coupled to node 1068) and a drain terminal 1124 coupled to a node 1128 (which is coupled to node 1060). Node 1128 is coupled to a drain terminal 1132 of a PMOS transistor 1136 and to a drain terminal 1140 of a PMOS transistor 1144. A gate terminal 1148 of PMOS transistor 1136 is coupled to node 1120, and a gate terminal 1152 of PMOS transistor 1144 is coupled to line 650. A source terminal 1156 of transistor 1136 and a source terminal 1160 of transistor 1144 are both coupled to V cc . Node 1128 is coupled to an input terminal of an inverter 1180 which has an output terminal coupled to bus 68 and to an input terminal of a 2-input NOR gate 1188. Similarly, node 1068 is coupled to an input terminal of an inverter 1192 which has an output terminal coupled to an input terminal of a 2-input NOR gate 1200. An output terminal of NOR gate 1188 is coupled to another input terminal of NOR gate 1200, and an output terminal of NOR gate 1200 is coupled to another input terminal of NOR gate 1188. NOR gates 1188 and 1200 thus function as a latch so that the signals at the output terminals of inverters 1180 and 1192 are maintained until the next comparison function. The output terminal of NOR gate 1200 is coupled to an input terminal of a 2-input NOR gate 1204. Another input terminal of NOR gate 1204 is coupled for receiving an active high DRAM RAS signal. The output terminal of NOR gate 1204 is coupled to SPUMP ( S low Pump) signal line 70 and to an input terminal of an inverter 1208. An output terminal of inverter 1208 is coupled to NSPUMP signal line 72. As previously discussed, node 650 from logic voltage level translator 40 in Fig. 7 switches between V cc -1.5 volts and V cc as node "W" switches between 0 volts V ss and +1.5 volts, respectively. Refer once again to the comparator 56 in Figure 8. Between sensing cycles node "W" is low, and NMOS transistor 1004 is off. At this time, signal 650 is also low and PMOS transistors 1076 and 1144 are on, charging nodes 1068 and 1128 up to V cc . No current is drawn during this time because there is no conductive path to V ss . When signal "W" transitions high (to +1.5 volts) node 650 also transitions high (to V cc ). NMOS transistors 1076 and 1144 turn off. As NMOS transistor 1004 turns on, node 1016 transitions low. If the translated substrate voltage (V bb + V cc ) on bus 58 is at a higher voltage than V ref , transistor 1032 starts to conduct before transistor 1024, since their source terminals are connected together. As transistor 1032 conducts, its drain 1104 discharges toward ground faster than the drain of transistor 1024 discharges toward ground. Preceding this, nodes 1068 and 1128 (the gate connections of transistors 1112 and 1052) were each at the same voltage, V cc . Thus, as the drain node 1104 of transistor 1032 discharges towards V ss , transistor 1112 conducts pulling node 1128 to ground. This turns off NMOS transistor 1052 preventing it from pulling node 1068 to ground while turning on PMOS transistor 1084 to maintain node 1068 at V cc . Thus under this condition with the translated substrate voltage (V bb + V cc ) on bus 58 above the reference voltage, node 1128 goes to ground while node 1068 remains at V cc . When node 1128 goes to ground, the output of inverter 1180 goes high (to V cc ). Note that after this initial switching transient but with nodes W and 650 still high, there is again no current path. There is no current in the right hand side of the circuit because PMOS transistors 1136 and 1144 are both off and there is no current in the left hand side because NMOS transistor 1052 is off. For as long as nodes "W" and 650 remain high, node 1128 remains low and node 68 remains high. Thus, a positive pulse on "W" (and on bus 650), when node 58 is above V ref , causes a positive pulse on node 68 while the output of inverter 1192 remains at ground. This positive pulse indicates pumping is required. (The translated substrate voltage is too positive.) This pulse does two things. First it provides a single pump cycle on node 68 delivered to the charge pump itself which will be described below. And second, it sets a flip flop comprised of NOR gates 1188 and 1200 into the appropriate state to insure the oscillator will operate at high frequency. The positive pulse on node 68 causes the output of NOR gate 1188 to go low. This low combined with the low out of inverter 1192 causes the output of NOR gate 1200 to go high. This high maintains the output of NOR gate 1188 low even after the positive pulse on node 68 terminates. Thus the output of NOR gate 1200 remains high (until such time that inverter 1192 provides a high output). If, instead, the translated substrate voltage (V bb + V cc ) on bus 58 is below the reference voltage, no pumping is required. In this case, node 1068 pulses to ground while node 1128 remains high. With node 1128 remaining high, there is no change in the low voltage on node 68, and no pump cycle is executed. At the same time, the positive pulse on the output of inverter 1192 causes NOR gate 1200 to have a low output. This low output combined with a low on node 68 causes a high output of NOR gate 1188. The high output of NOR gate 1188 maintains a low on the output of NOR gate 1200 even after the positive pulse on the output of inverter 1192 terminates. Thus, if the comparator most recently determined that the substrate was too positive (i.e., pumping is required) the output of NOR gate 1200 will be high. If, instead, the comparator most recently determined that the substrate was sufficiently negative (i.e, no pumping required), the output of NOR gate 1200 will be low. The signal RASD is high whenever the circuit is in its active state. For a DRAM, this would be high during an active cycle and low during precharge between cycles. When either the comparator most recently determined that pumping is required or when the circuit is active (RASD is high) then the output of NOR gate 1204 (SPUMP) is low. A low output on SPUMP ( S low Pump) causes the oscillator not to oscillate slow but rather fast. As previously discussed, when SPUMP is low , the oscillator operates at high frequency which in turn permits high pumping current. When SPUMP is high , the oscillator operates at a much lower frequency, as does the logic voltage level translator 40 of Figure 7, the substrate voltage translator 44 of Figure 8 and the comparator 56 of Figure 8. The low frequency operation of approximately 200 kilohertz, with much of the circuitry operating from a 1.5 volt supply, permits the pump to consume less than 1 microamp of total current when no pumping is required. Yet when pumping is required, the circuit automatically switches to high frequency, and at high frequency is capable of pumping more than 1 milliamp of current out of the substrate. In conclusion, the present invention translates low voltage swing logic nodes to high voltage swing nodes without any power-consuming direct current paths. This in turn can be used to translate a negative substrate voltage to a positive voltage to provide for easy comparison to a reference voltage. When applied in a low-power charge pump, the comparison provides the ability to turn the pomp off when pumping is not needed, thereby providing a pump with a reduced-power standby mode. In addition, the ability of the voltage translation circuit to operate without adding substrate current further reduces the power consumption of the charge pump. While the above is a complete description of specific embodiments of the present invention, various modifications may be employed. For example, assumed voltages and the sizes of the various transistors may vary without departing from the principles of operation.

Claims: An apparatus for translating voltages comprising: a first capacitance (450) having a first terminal coupled to receiving a first control signal (X), and a second capacitance (452) having a first terminal coupled for receiving a second control signal (Y); a first transistor (476) and a second transistor (460) having their respective gates and first current flowing terminals cross-coupled to second terminals (461, 463) of said first and second capacitance (450, 452), respectively; wherein each of said control signals (X, Y) comprises a low portion and a high portion, and wherein either the low portion of the first signal is mutually exclusive with the low portion of the second signal or the high portion of the first signal is mutually exclusive with the high portion of the second signal; characterized in that said apparatus further comprises: a third transistor (660) having a first current flowing terminal coupled to a high voltage current source, a control terminal coupled to the second terminal (463) of the first capacitance (450), and a second current flowing terminal; a fourth transistor (680) having a first current flowing terminal coupled to the second current flowing terminal of the third transistor (660) at a first node (672), a second current flowing terminal coupled to a ground potential (Vss), and a control terminal coupled to the first terminal of the first capacitance (450); wherein one of the third or fourth transistor (660, 680) conducts when the first signal is received by the first capacitance (450); and wherein the other of the third or fourth transistor (660, 680) conducts when the second signal is received by the second capacitance (452). The apparatus according to claim 1, wherein the third transistor (660) comprises a PMOS transistor and the fourth transistor (680) comprises a NMOS transistor. The apparatus according to claim 1 or 2 wherein said apparatus is used to translate a substrate voltage, said apparatus further comprising: a fifth transistor (710) having a first current flowing terminal coupled to the high voltage current source, a control terminal coupled to the second terminal (463) of the second capacitance (452), and a second current flowing terminal; a sixth transistor (730) having a first current flowing terminal coupled to the second current flowing terminal of the fifth transistor (710) at a second node (720), a second current flowing terminal coupled to a ground potential (Vss), and a control terminal coupled to the first terminal of the second capacitance (452); wherein one of the fifth or sixth transistor (710, 730) conducts when the first control signal is received by the first capacitance (450); wherein the other of the fifth or sixth transistor (710, 730) conducts when the second control signal is received by the second capacitance (452); a third capacitance (820) having a first terminal coupled to the first node (672); a fourth capacitance (812) having a first terminal coupled to the second node (720); a seventh transistor (854) having a first current flowing terminal coupled to a second terminal of the third capacitance (820), a second current flowing terminal coupled to the substrate (Vbb), and a control terminal coupled to a second terminal of the fourth capacitance (812); an eighth transistor (830) having a first current flowing terminal coupled to the second terminal of the fourth capacitance (812), a second current flowing terminal coupled to the substrate (Vbb), and a control terminal coupled to the second terminal of the third capacitance (820); wherein the seventh transistor (854) conducts current when the fifth transistor (710) conducts; wherein the eighth transistor (830) conducts when the third transistor (660) conducts; wherein a voltage at the second terminal of the fourth capacitance (812) is approximately equal to the substrate voltage when the eighth transistor (830) conducts; and wherein a voltage at the second terminal of the fourth capacitance (812) is approximately equal to the substrate voltage plus the voltage of the high voltage current source when the seventh transistor (854) conducts. The apparatus according to claim 3 further comprising: a fifth capacitance (804) having a first terminal coupled to the first terminal of the fourth capacitance (812); and a ninth transistor (904) having a first current flowing terminal coupled to a second terminal of the fifth capacitance (804) at an output node (58), a second current flowing terminal coupled to the substrate (Vbb), and a control terminal coupled to the control terminal of the eighth transistor (830); and wherein the ninth transistor (904) conducts current when the third transistor (660) conducts current for causing the output node (58) to be at the substrate voltage. The apparatus according to claim 4 further comprising: a tenth transistor (880) disposed between the second terminal of the fifth capacitance (804) and the first current flowing terminal of the ninth transistor (904), said tenth transistor (880) having a first current flowing terminal coupled to the second input terminal of the fifth capacitance (804) at the output node (58), a second current flowing terminal coupled to the first current flowing terminal of the ninth transistor (904), and a control terminal coupled to the control terminal of the ninth transistor (904); and an eleventh transistor (930) having a first current flowing terminal coupled to the high voltage current source, a second current flowing terminal coupled to the first current flowing terminal of the ninth transistor (904), and a control terminal coupled to the second terminal of the fifth capacitance (804). The apparatus according to any one of the claims 1 to 5 wherein the first and second transistors (476, 460) each comprise PMOS transistors. The apparatus according to any one of the claims 3 to 6 wherein the third and fifth transistors (660, 710) each comprise a PMOS transistor and the fourth and sixth transistors (680, 710) each comprise a NMOS transistor. The apparatus according to any one of the claims 3 to 7 wherein the seventh, eighth, ninth, tenth, and eleventh transistors (854, 830, 904, 880, 930) each comprise a NMOS transistor. The apparatus according to any one of the claims 1 to 8 wherein the low portion of the first and second control signals are each approximately zero volts and wherein the high portion of the first and second control signals are each approximately 1.5 volts. The apparatus according to any one of the claims 1 to 9 wherein the high voltage current source is approximately five volts.

Inventors: PROEBSTING ROBERT J, PROEBSTING, ROBERT J.
Applicants: HYNIX SEMICONDUCTOR INC, HYNIX SEMICONDUCTOR INC.
Application year: 1997
Main classification: G05F3
IPC codes: G11C5, G05F3, H03B5, H02M3