Frequency-to-current converter

Abstract: A frequency-to-current converter (10) includes several capacitances with capacitive values that are effectively multiplied. After each of a series of periodic pulses, the voltage on a "ramp" capacitance (C ramp ) is charged to a starting voltage (V start ). Then, during the period preceding the subsequent pulse, the ramp capacitance is allowed to discharge at a discharge rate that is a function of a voltage (VCCB) on a discharge-current bias capacitance (CCB). At the end of the period, the voltage on the ramp capacitance is sampled and compared to a reference (V fixed ). If the voltage (V ramp ) on the ramp capacitance (C ramp ) is too low or too high, indicating a discharge current that is too high or too low, respectively, the bias voltage (VCCB) on the bias capacitance (CCB) is adjusted to compensate for the error. In another embodiment, a small ramp capacitance is repetitively charged and discharged between two reference voltage levels using alternating charge and discharge current levels. The charge and discharge current levels are related to output current levels so that the rate at which the ramp capacitance transitions between the two reference voltages provides an indication of the output current.

Description: BACKGROUND Field of the Invention This invention relates generally to frequency-to-voltage and frequency-to-current converters. Description of Related Technology Frequency-to-current converters are typically implemented by combining a frequency-to-voltage converter and a voltage-to-current converter. In a conventional frequency-to-voltage converter, a one-shot-controlled current reference is averaged by an integrating amplifier. The one-shot and the integrating amplifier each employ rather large capacitances that are difficult to integrate, and are therefore typically provided externally. The remaining circuitry may then be provided as one or more integrated circuits. Many conventional frequency-to-voltage converters are available, several of which are described in the 1994 Applications Handbook available from Burr-Brown of Tucson, Arizona. An additional problem of conventional frequency-to-voltage converters results from a trade-off between ripple and settling time. High-resolution frequency-to-voltage converters demand low ripple, but decreasing trade-off between settling time and ripple can be improved by filtering the frequency-to-voltage output with a low-pass filter. Unfortunately, such low-pass filters require some additional capacitance, making the frequency-to-voltage converter even more difficult to integrate. Finally, conventional frequency-to-current converters are typically open-loop systems. As such, conventional frequency-to-current converters are sensitive to external disturbances to the output, such as leakage current. Because of the foregoing problems, there exists a need for a frequency-to-current converter that is insensitive to output disturbances, relies on matched component values to ensure accuracy, and makes use of relatively small, integrable, capacitances. U. Tietze, Ch. Schenk: "Halbleiterschaltungstechnik", 6. Aufl., 1985, Springer, Berlin, pages 183, 462/463 discloses a function generator. DE-A-2 117 599 describes a method and a device for converting a variable input frequency into a proportional output voltage. FR-A-2 281 677 reveals a frequency voltage converter. SUMMARY The present invention addresses the aforementioned needs by providing a closed-loop frequency-to-current converter with integrable capacitances. To accomplish this, a frequency-to-current converter in accordance with the present invention includes several capacitances with capacitive values that are effectively multiplied using inventive capacitance-multiplication techniques. One method in accordance with the invention allows for the conversion of a periodic series of input pulses to a proportional current without the use of an external capacitance. After each incoming pulse, the voltage on a "ramp" capacitance is reset to a starting voltage. Then, during the period preceding the subsequent pulse, the ramp capacitance is allowed to discharge at a rate that is controlled by a voltage stored across a discharge-current bias capacitance. Next, following the subsequent input pulse, the voltage on the ramp capacitance is sampled for a sample period that is short relative to the period of the pulse train. During the sample period, the voltage on the ramp capacitance is compared to a fixed reference voltage. If the voltage on the ramp capacitance is lower than the fixed reference voltage, the discharge rate (and therefore the output current) is too high. Conversely, if the voltage on ramp capacitance is higher than the fixed reference voltage, the discharge rate is too low. When the discharge rate of the ramp capacitance is determined to be either too low or too high, the stored voltage across the discharge-current bias capacitance is adjusted to compensate for the error. For example, if after one period of the input signal the voltage on the ramp capacitance is too high, the voltage on the bias capacitance is elevated to increase the discharge rate of the ramp capacitance for the next period. Once the comparison has been made and the voltage on the bias capacitance adjusted accordingly, the ramp capacitance is reset to the starting voltage and the cycle begins again. As will be explained in detail below, the sampling technique is beneficial because the technique 1. allows for the use of a much smaller bias capacitance than would otherwise be required to provide a desired time constant, 2. allows a user to change the time constant of the bias capacitance without changing the value of the capacitance, and 3. eliminates the need to adjust the value of the bias capacitance for different input frequencies. These benefits combined allow the bias capacitance to be integrated. Another method in accordance with the invention, hereafter referred to as a "folding" technique, allows for the use of a relatively small ramp capacitance to establish the appropriate output current level. In accordance with this method, the small ramp capacitance is repetitively charged and discharged between two reference voltage levels using alternating charge and discharge current levels, each of which is a function of the output current level. Because the charge and discharge current levels are related to the output current level, the rate at which the ramp capacitance is charged and discharged between the two reference voltages provides an indication of the output current. For example, if the output current is too high, the related charge and discharge currents will also be too high, resulting in relatively rapid charge and discharge rates. And, due to the rapid charge and discharge rates, the number of times the ramp capacitance will charge and discharge over a given period of time will increase in proportion to the increase in output current. Thus, a method in accordance with the invention includes the step of counting the number of times the ramp capacitance charges and discharges over a given period of time to determine whether to increase or decrease the output current. The inventive folding technique allows for a significant reduction in the size of the ramp capacitance, thereby allowing the ramp capacitance to be easily integrated. Thus, in accordance with the present invention, both the bias capacitance and the ramp capacitance may be advantageously integrated with the other components of the frequency-to-current converter. Moreover, the inventive frequency-to-current converter design takes advantage of the exceptional device matching obtained using integrated circuit technology, as opposed to exact device values, to ensure accurate frequency-to-current conversion. BRIEF DESCRIPTION OF THE FIGURES These and other features, aspects, and advantages of the present invention will become understood with regard to the following description, appended claims, and accompanying drawings, where: Figure 1 shows a frequency-to-current converter 10 in accordance with one embodiment of the present invention; Figure 2 shows various waveforms associated with frequency-to-current converter 10 of Figure 1; Figure 3 shows a frequency-to-current converter 50 in accordance with another embodiment of the present invention; Figure 4 illustrates the concept of "folding" the voltage range applied across a capacitance; Figure 5 is a schematic diagram of one embodiment of folded ramp capacitance 60; Figure 6 shows various waveforms associated with a "folded" capacitance; Figure 7 shows various waveforms associated with folded ramp capacitance 60; Figure 8 shows a frequency-to-current converter 150 in accordance with yet another embodiment of the present invention; and Figure 9 is a schematic diagram of a folded ramp capacitance 160. DETAILED DESCRIPTION Figure 1 shows a frequency-to-current converter 10 in accordance with one embodiment of the present invention. Frequency-to-current converter 10 includes a sample timer 12, a ramp circuit 20, a differential transconductance amplifier GM1, a first sample gate 30, a second sample gate 31, and an output current generator 32. Sample timer 12 is connected to the gate of a transistor 22 within ramp circuit 20 via a reset line RESET, to the control gate of sample gate 30 via a line SAMPLE, and to the control gate of sample gate 31 via a line SAMPLE . Depending on the voltage applied from sample timer 12 to the control terminal of transistor 22, transistor 22 transfers a reference voltage on a terminal V START to a terminal of a ramp capacitance C RAMP within ramp circuit 20. The non-inverting (+) input of transconductance amplifier GM1 is connected to a line V RAMP shared by transistor 22, capacitance C RAMP , and through sample gate 31 to an output of current generator 32. The inverting (-) terminal of transconductance amplifier GM1 is connected to a fixed reference voltage on terminal V FIXED , and the output of transconductance amplifier GM1 is connected to a current-handling terminal of sample gate 30. The other current-handling terminal of sample gate 30 is connected to the control gates of transistors 34 and 36 in current generator 32, and to one terminal of a discharge-current bias capacitance C CB . Current generator 32 provides an output current I OUT that is proportional to the input frequency of an input signal on a terminal F INPUT , and is therefore inversely proportional to the average period of the input signal on terminal F INPUT . Stated mathematically, I OUT = α T INPUT where T INPUT is the average period of e.g. one hundred cycles of the input signal in terminal F INPUT and α is the gain factor of frequency-to-current converter 10. In the circuit of Figure 1, output current I OUT is negative (i.e., the current flows into current generator 32). Of course, current generator 32 may also be configured (e.g., using a conventional current mirror) to provide a positive output current I OUT . Conventional current sources 35 and 37 provide offset currents through transistors 34 and 36, respectively, so that frequency-to-current converter 10 provides some specified offset current when the signal on terminal F INPUT has a frequency of between zero Hz and some selected value. Current sources 35 and 37 each source the same level of current so that the ramp current I RAMP is equal to the output current I OUT . Where desired, the offset current can be eliminated or set to zero. Figure 2 shows various waveforms associated with frequency-to-current converter 10 of Figure 1. The timing of the waveforms of Figure 2 is not to scale. For example, the period T INPUT of the input signal on terminal F INPUT is, in one embodiment, 300us as compared to the reset period T RES of 1us and the sample period T SAM of from 100ns to 800ns. Sample timer 12 is a conventional timing circuit that, based on its function described herein, may easily be implemented by those of skill in the art. Referring to the waveforms of Figure 2 in conjunction with frequency-to-current converter 10 of Figure 1, sample timer 12 outputs a sample pulse of period T SAM on line SAMPLE in response to an incoming pulse on terminal F INPUT . The sample pulse on line SAMPLE turns on sample gate 30, thereby connecting the output of transconductance amplifier GM1 to a terminal of bias capacitance C CB . During the sample period T SAM , a logic zero on line SAMPLE turns gate 31 off so that the voltage on terminal V RAMP remains constant from e.g. time t 1 to time t 2 . During the sample period T SAM between times t 1 and t 2 the voltage on terminal V RAMP is less than the voltage on terminal V FIXED . Thus, transconductance amplifier GM1 will draw charge away from capacitance C CB , thereby decreasing the voltage on the gates of transistors 34 and 36. As a consequence, the current I RAMP through transistor 34 and the output current I OUT through transistor 36 will decrease. Subsequent to the sample pulse, sample timer 12 outputs a reset pulse of period T RES to the gate of transistor 22 in ramp circuit 20. The reset pulse turns transistor 22 on, charging C RAMP to the reference voltage level on terminal V START , as shown in Figure 2 at time t 3 . Once the voltage on terminal V RAMP (and consequently across ramp capacitance C RAMP ) has been reset to the reference voltage on terminal V START , ramp capacitance C RAMP will discharge through transistor 34 so that the voltage level on terminal V RAMP will decrease to a level V FIN at time t 4 when the next input pulse arrives on line 42. Because the voltage V FIN on line V RAMP is below the voltage on terminal V FIXED , during the sample period from t 4 to t 5 transconductance amplifier GM1 will again draw charge away from bias capacitance C CB , thereby decreasing the voltage on terminal V CCB . This effect is shown in Figure 2 as the decrease in the voltage V CCB that occurs during the second sample period (from t 4 and t 5 ). Having decreased the bias voltage on terminal V CCB , the current through transistors 34 and 36 will also decrease. Thus, capacitance C RAMP will discharge more slowly than during the prior period, as evidenced by the decreased slope of the voltage on terminal V RAMP from time t 6 to time t 8 . When, as in the third sample period of Figure 2 (from time t 7 to time t 8 ), the finish voltage V FIN on ramp capacitance C RAMP is greater than the voltage on terminal V FIXED , transconductance amplifier GM1 will provide charge to bias capacitance C CB through sample gate 30 during the sample period T SAM , as evidenced by the voltage increase on terminal V CCB from time t 7 to time t 8 . The above-described closed-loop feedback system will eventually cause the voltage on terminal V RAMP to go from the reference voltage on terminal V START to the reference voltage on terminal V FIXED in the period between input pulses. In other words, the voltage level V FIN for the voltage on terminal on V RAMP will equal the voltage level on terminal V FIXED during a sample period T SAM . In such a case, virtually zero current will flow from the output of transconductance amplifier GM1 during the sample period T SAM . This case is illustrated in Figure 2 between time t 10 and time t 11 . It is important here, as with conventional feedback systems, that the time constant τ Ccb of current-bias capacitance C CB be selected to provide a fast response without resulting in instability. The time constant τ Ccb (measured in units of time) may be calculated as: τ Ccb = C Ccb g mCcb where C Ccb is the capacitance of current-bias capacitance C CB and g mCcb is the transconductance providing the charge/discharge current for C CB . The time constant τ Ccb must be large enough to provide the appropriate response. The time constant τ Ccb may be increased by decreasing the value of the transconductance g m1 of transconductance amplifier GM1, which provides the charge and discharge currents to capacitance C CB . However, there is a lower limit of g ml below which noise becomes unacceptably high. Another possibility is to increase the value of capacitive C Ccb . Unfortunately, the frequencies of interest (e.g., 1/T INPUT or 3.3KHz) require a relatively large capacitance that may not be integrated practically. The present invention overcomes the foregoing problems associated with increasing the time constant τ Ccb by employing an inventive sample-and-hold technique. Sample timer 12 and sample gate 30 sample and hold the output of transconductance amplifier GM1 so that the transconductance g mCcb input to capacitance C CB is reduced to g mCcb = g m1 × T SAM T INPUT Thus, the time constant of C CB , τ Ccb , is increased to: τ Ccb = C Ccb ( T SAM T INPUT ) g m1 or τ Ccb = C Ccb g m1 × T INPUT T SAM As equation (7) makes clear, the "effective" capacitance of C Ccb may be expressed as C eff = C Ccb (T INPUT /T SAM ). In other words, the necessary value (and therefore size) of current-bias capacitance C CB is reduced by a factor of T SAM /T INPUT ; accordingly, C CB may be reduced to a integrable size while maintaining the appropriate time constant τ Ccb . Without the inventive sample-and-hold technique, the value of C CB would have to be adjustable to allow for different input frequencies. Fortunately, the inventive sample-and-hold technique eliminates the need to provide a variable capacitance. Substituting 1/T INPUT in equation (6) with the input frequency f INPUT shows that the time constant τ Ccb of bias capacitance C CB is a function of the input frequency f INPUT . Stated mathematically, τ Ccb. = 1 f INPUT × C Ccb T SAM g m1 Thus, using the sample-and-hold technique that samples for a fixed time period T SAM and a fixed capacitance C ccb , the time constant τ Ccb automatically changes with changes in the input frequency f INPUT . Figure 3 shows a frequency-to-current converter 50 in accordance with another embodiment of the present invention. To the extent that Figure 3 is similar to Figure 1, similar reference numbers are used. Frequency-to-current converter 50 functions in substantially the same way as described above in connection with Figures 1 and 2. However, in frequency-to-current converter 50, ramp capacitance C RAMP is replaced by a folded ramp capacitance 60. For simplicity, the optional offset current sources 35 and 37 are not shown in Figure 3. In accordance with the embodiment of Figure 3, ramp capacitance C RAMP is replaced by a folded ramp capacitance 60. As explained below, the process of "folding" allows for the use of a relatively small value of ramp capacitance C RAMP , resulting in easier, more economical, integration. Folded ramp capacitance 60 receives as inputs a reference voltage on terminal V MAX and a second reference voltage on a terminal V MIN , the voltages on terminals V MAX and V MIN being e.g. 5 volts and 1 volt, respectively. In one embodiment, terminals V MAX and terminal V REF are connected so that the reference voltage on terminal V REF also serves as the reference voltage on terminal V MAX . Capacitance 60 outputs a signal "UNDER" on a line 62 to an input of a transconductance amplifier GM2. Transconductance amplifier GM2 is configured to operate as a conventional differential amplifier when a logic zero (e.g., zero volts) is applied on line 62, and to output its maximum positive output current, regardless of the levels on its differential input terminals (+ and -), when a logic one (e.g., 5 volts) is applied on line 62. The following explanation describes the concept of "folding" in general, which may be used to increase the time constant of a given capacitance C. In the general case, for a given capacitance C with a constant charging current I C , the time constant τ C is proportional to the product of the change in voltage ΔV and the capacitance C divided by the current I C . This relationship may be expressed mathematically as: τ C = CΔV I C Thus, for a given time constant τ C , the necessary value of the capacitance C, and thus the physical area required to integrate such a capacitance, may be minimized by either increasing ΔV or decreasing the current I C . Unfortunately, there is a lower limit to the level of current I C used to charge the capacitance C, for when the current is too low, noise and leakage produce unacceptably poor accuracy in defining the time constant τ C . Moreover, it is not practical to substantially increase the power supply voltages for integrated circuits to increase the effect of capacitance of a particular capacitance on the integrated circuit. For these reasons, Applicant invented a folding technique that allows for an effective increase in the voltage change ΔV across ramp capacitance C RAMP without requiring increased voltage potentials. According to this aspect of the invention, the desired voltage range is "folded" into a number of smaller voltage ranges. Figure 4 illustrates the concept of "folding" the voltage range applied across a capacitance. Paths P 1 and P 2 are discharge paths similar to the V RAMP discharge paths between times t 3 and t 4 , between times t 6 and t 7 , and between times t 9 and t 10 of Figure 2. Path P 1 shows the voltage across capacitance C discharging at a constant current I C from 5 volts to -11 volts (for a total ΔV of 16 volts) over a time period T. Path P 2 shows the voltage across capacitance C discharging at a reduced constant current I C ' so that during the period T the capacitance C discharges from 5 volts to -9 volts, for a total ΔV of only 14 volts. Assuming that the current used to produce path P 1 is the desired current, the voltage difference between path P 1 and path P 2 at the end of period T (i.e., the error voltage V ε of 2 volts) may then be used to provide feedback to transconductance amplifier GM2 in the manner described above in connection with transconductance amplifier GM1. According to the inventive folding technique, the 16-volt range of path P 1 may be folded into a number of segments. For example, Figure 4 shows path P 1 folded into a path P 1 ', which includes four segments, two decreasing and two increasing. The folded path P 1 ' represents a voltage change ΔV having an absolute value of 16 volts (i.e., four 4-volt changes) without exceeding 5 volts or going below 1 volt. Similar to path P 1 , path P 2 may folded into e.g. four segments, two decreasing and two increasing, as shown using path P 2 '. Importantly, at the end of period T the folded waveform P 2 ' is off by an error voltage V εF of the same magnitude as the error voltage V ε separating paths P 1 and P 2 . The error voltage V εF may be used to provide feedback for transconductance amplifier GM2. Note that the error voltage V εF is of the opposite polarity of the voltage V ε . For this reason, the inverting and non-inverting (- and +) terminals of transconductance amplifier GM2 are reversed in the embodiment of Figures 3 and 5 as compared to the terminals of transconductance amplified GM1 of Figure 1. This and other aspects of the embodiment of Figure 3 are described below in detail in connection with Figures 5 and 6. Figure 5 is a schematic diagram of one embodiment of folded ramp capacitance 60. Folded ramp capacitance 60 includes ramp capacitance C RAMP , a comparator circuit 70, a counter 80, and a current source 90. Current source 90 receives the ramp current I RAMP on line 92 from current generator 32 through sample gate 31. The ramp current I RAMP is mirrored by transistors 94 and 96 to provide a current I RAMP ' substantially equal to the ramp current IRAMP through series-connected transistors 98 and 100. Transistors 102 and 104 conventionally mirror transistors 98 and 100 so that the current I RAMP ' travels through series-connected transistors 96, 98, and 100, and through series-connected transistors 106, 102, and 104. A transistor 108 mirrors transistor 106 so that a current I C substantially equal to the current I RAMP ' through transistor 106 is conducted by the current-handling terminals of transistor 108. Finally, a current source 110, which includes a transistor 112 and a transistor 114, is biased by the combination of transistors 102 and 104. The pair of transistors 112 and 114 is similar to transistors 102 and 104, except that transistors 112 and 114 have a multiplication factor M of two. Because of the multiplication factor, current source 110 conducts twice as much current for a given bias voltage on the base of transistor 112 as the series connected transistors 102 and 104 (and consequently twice as much current as the current I C through transistor 108). With this configuration, when transistor 114 is off, the folded ramp current I F from current source 90 is substantially equal to I C . Conversely, when transistor 114 is conducting, current source 110 conducts a current substantially equal to twice I C so that the folded ramp current I F is equal to I C - 2I C = -I C . Thus, the folded ramp current I F may be used to alternately charge and discharge ramp capacitance C RAMP at rates of I C and -I C , respectively, the direction of folded ramp current I F depending on whether transistor 114 is on or off. Comparator circuit 70 receives the folded ramp current I F on an input line 71. Referring to Figures 4 and 5, path P 1 ' will be used to explain the operation of comparator circuit 70, where the path P 1 ' represents the changing voltage level on terminal V RAMP of Figures 3 and 5. Referring to Figure 5, if we assume that the reset terminal RESET of counter 80 was just supplied a positive (e.g., 12-volt) pulse, the voltage on terminal V RAMP will be reset to 5 volts, as shown at the Y axis of Figure 4. And, if we assume that the folded ramp current I F from current source 90 is equal to -I C (i.e., that transistor 114 is biased on), the voltage on terminal V RAMP will begin to discharge as shown in Figure 4 as element A. Comparator circuit 70 includes a maximum-voltage comparator 72, a minimum-voltage comparator 74, a reset transistor 76, and a conventional cross-coupled latch 78. When the voltage on terminal V RAMP on the ramp capacitance C RAMP discharges below V MIN , comparator 74 will output a logic one to latch 78, which will consequently output a logic zero to an input of an AND gate 84. For the time being, assume that other input to AND gate 84 from line 62 is a logic one, so that the logic zero from latch 78 will cause AND gate 84 to output a logic zero to the gate of transistor 114, turning transistor 114 off. With transistor 114 off, the folded ramp current I F from current source 90 will change from -I C to +I C , the current through transistor 108. As shown in Figure 4 as leg B of path P 1 ', the folded ramp current I F will begin to charge the ramp capacitance C RAMP , thereby increasing the voltage on terminal V RAMP . Next, when the voltage on terminal V RAMP exceeds the voltage on terminal V MAX (5 volts in this case), comparator 72 will output a logic one to latch 78 causing latch 78 to output a logic one to transistor 114. As a result, transistor 114 and transistor 112 will conduct twice as much current (2I C ) as transistor 108 (I C ), so that the folded ramp current I F of current source 90 will return to -I C . Ramp capacitance C RAMP will consequently begin to discharge, resulting in the steady decrease in the voltage on terminal V RAMP shown as leg C of path P 1 ' in Figure 4. As long as there is an ramp current I RAMP through line 92, there will be a folded ramp current I F that is either charging or discharging ramp capacitance C RAMP . In the case of path P 1 ', the voltage on terminal V RAMP of ramp capacitance C RAMP is shown to discharge to 1 volt twice (legs A and C) and charge to 5 volts twice (legs B and D). Counter 80 is connected to comparator circuit 70 on line UNDER_L, which stands for "under low," indicating that the voltage on terminal V RAMP is under the minimum voltage on terminal V MIN . The under-low pulses on line UNDER_L to counter 80 allow counter 80 to count the number of times the changing ramp voltage on terminal V RAMP transitions from negative-going to positive-going, thereby providing an indication of the number of "folds." Once the count received by counter 80 goes above a predetermined number, counter 80 outputs a logic zero on output line 62 to transconductance amplifier GM2 and to AND gate 84. Figure 6 is similar to Figure 4 in that Figure 6 illustrates an example where the voltage across ramp capacitance C RAMP is folded four times. The voltage V RAMP on ramp capacitance C RAMP transitions from 5 volts to 1 volt and then from 1 volt to 5 volts as described above in connection with Figure 4. Each time the voltage transitions from negative-going to positive-going, comparator 74 of comparator circuit 70 outputs a positive pulse on line UNDER_L. This positive pulse increments the count stored in counter 80. In the example of Figure 6, counter 80 is configured to output a logic zero on line 62 only after the count is equal to two. In other words, as long as the count is "under" two, counter 80 will output a logic one as the signal UNDER on line 62. If the ramp current I RAMP is far too low for a given input frequency, the folded ramp current I F of current source 90 will also be too low. In such a case, counter 80 will not reach a count of two during one period of the input signal. In such a case, counter 80 will output a logic one on line 62, which will deactivate GM2 so that GM2 will not respond to its positive and negative inputs, but will instead output its maximum output current of, for example, 10uA. Thus, when the output current I OUT is far low than is appropriate for a given input frequency, transconductance amplifier GM2 outputs a maximum current to charge bias capacitance C CB . As the value of the output current I OUT (and therefore the ramp current I RAMP ) increases to more accurately reflect the input frequency, the transition time of the ramp voltage on terminal V RAMP will continue to decrease until the counter reaches a count of two prior to the end of one period T INPUT of the input signal. This case is shown in Figure 6. Once counter 80 reaches a count of two, counter 80 outputs a zero on line 62, which activates transconductance amplifier GM2. Transconductance amplifier GM2 than responds to its inverting and non-inverting inputs. The active region of transconductance amplifier GM2 is illustrated in Figure 6 as "GM2 active." Once GM2 is active, frequency-to-current converter 50 works in substantially the same way as described above in connection with frequency-to-converter 10. If at the end of the period T INPUT of the input signal the ramp voltage on terminal V RAMP is higher than the reference voltage on terminal V MAX , transconductance amplifier GM2 draws some charge from bias capacitance C CB during the sample period T SAM , thus decreasing the ramp current I RAMP (and consequently decreasing the folded ramp current I F ). Conversely, if GM2 is in the active region and the ramp voltage on terminal V RAMP is below the reference voltage on terminal V MAX at the end of the period T INPUT , then the active transconductance amplifier GM2 will output current to charge bias capacitance C CB through sample gate 30 during the sample period T SAM . Using a folding factor of four, the embodiment described in connection with Figure 6 increases the effective capacitance of ramp capacitance C RAMP by a factor of four. Of course, the voltage across ramp capacitance C RAMP can be folded any number of times. For example, Figure 7 shows various waveforms associated with folded ramp capacitance 60 in an embodiment that multiplies the total voltage change across ramp capacitance C RAMP by thirty-two times. During the sample period T SAM of Figure 7, the voltage on terminal V RAMP is slightly less than the reference voltage on terminal V MAX . As a result, transconductance amplifier GM2 provides an output current to bias capacitance C CB , thereby increasing the bias voltage on terminal V CCB . Then, during the subsequent reset pulse, the positive reset voltage on the gate of transistor 76 of comparator circuit 70 (see Figure 5) resets the voltage on terminal V RAMP to the voltage on terminal V MAX . Once reset, the voltage V RAMP transitions back and fourth between the voltage on terminal V MAX and the voltage on terminal V MIN approximately 32 times before the next sample pulse. In the case of Figure 7, counter 80 would be configured to activate GM2 after a count of 16. If counter 80 did not sense 16 negative to positive transitions between input pulses, counter 80 would output a logic one on line 62, indicating that the count was "under," causing transconductance amplifier GM2 to output the maximum available current to bias capacitance C CB . If the folded ramp current I F were far too high relative to the input frequency, the transitions of the ramp voltage on terminal V RAMP would occur far more rapidly, causing counter 80 to reach its maximum count and activate transconductance amplifier GM2 well before the end of the period T INPUT . When this happens, the signal from counter 80 on line 62 goes low, causing AND gate 84 to disable transistor 114, consequently disabling the folding action of folded ramp capacitance 60. As a result, the voltage on terminal V RAMP will continue to climb beyond the reference voltage on terminal V MAX (to a maximum voltage of e.g. 4 volts above V MAX , or 9 volts), as shown in Figure 6. Transconductance amplifier GM2 will then output a current to decrease the ramp current I RAMP in much the same way as described above in connection with Figure 3. After a number of periods of the input signal, the final voltage on terminal of V RAMP after a single input period T INPUT will be precisely equal to the voltage on terminal V MAX . Figure 8 is a schematic diagram of yet another embodiment of the present invention -- frequency-to-current converter 150 -- wherein the settling time of the output current I OUT is improved by the addition of a variable current generator 200 and associated circuitry. The schematic of Figure 8 is similar to that of Figure 3, like elements having the same reference numbers. A folded ramp capacitance 160 is connected to a transconductance amplifier GM3 via a GM-enable line GM_EN and to current generator 200 via a current-select bus I SEL , which includes lines CNT<8, 8≤CNT<12, 12≤CNT<16, 17<CNT≤20, and CNT>20 (shown in Figure 9). Figure 9 is a schematic diagram of a folded ramp capacitance 160 that is similar to folded ramp capacitance 60 of Figure 5, like elements having the same reference numbers. When the count stored in a counter 180 is less than eight, counter 180 outputs a logic one on line CNT<8 of bus I SEL . Similarly, when the stored count is from eight to eleven, twelve to fifteen, eighteen to twenty, or greater than twenty-one, counter 180 outputs a logic one on the corresponding one of lines 8≤CNT<12, 12≤CNT<16, 17<CNT≤20, and CNT>20, respectively. When the stored count is sixteen or seventeen, a logic one on line GM_EN enables transconductance amplifier GM3 to function in the active mode, as described above for transconductance amplifier GM2 in connection with Figure 6. Current generator 200 of Figure 8 is conventionally configured to output a current of: 1. 1mA when the voltage on line CNT<8 represents a logic one; 2. 100µA when the voltage on line 8≤CNT<12 represents a logic one; 3. 10µA when the voltage on line 12≤CNT<16 represents a logic one; 4. -100µA when the voltage on line 17<CNT≤20 represents a logic one; and 5. -1mA when the voltage on line CNT>20 represents a logic one. Thus configured, the current used to charge and discharge bias capacitance C CB in frequency-to-current converter 150 varies depending on the count stored in counter 180. That is, when the voltage on terminal V CCB is relatively far from that required to provide the correct output current I OUT , the charge or discharge current used to correct the voltage on terminal V CCB is increased. Then, as the voltage on terminal V CCB approaches the appropriate fixed voltage V FIXED during the sample period T SAM , the charge or discharge current decreases. In this way, frequency-to-current converter 150 improves the settling time of ramp current I RAMP while maintaining stability. Frequency-to-current converters in accordance with the present invention are not limited to the particular applications described above. For example, in one embodiment, sample gate 31 of Figure 3 is eliminated by connecting line SAMPLE directly to the gate of transistor 104 and, through an additional input (not shown) of AND gate 84, to the gate of transistor 114. Furthermore, frequency-to-current converters that do not require the level of precision provided by the "active" analog mode of the second embodiment could rely only on the count stored in counter 80 to determine the output current level. Therefore, the scope of the appended claims should not be limited to the description of the preferred versions described herein.

Claims: A frequency-to-current converter comprising: an input node for receiving an input signal having an input frequency; a sample timer (12) connected to the input node, the sample timer for providing a sample signal on a sample terminal and a reset signal on a reset terminal; a ramp capacitance circuit (20) having a ramp capacitance, the ramp capacitance including a first terminal and a second terminal, the second terminal switchably connected to a first voltage reference in response to the reset signal from the sample timer; a transconductance amplifier (GM1) having an input terminal connected to the second terminal of the ramp capacitance and an output terminal, the transconductance amplifier providing an output current on the output terminal; a sample gate (30) having a first current-handling terminal connected to the output terminal of the transconductance amplifier, a control terminal connected to the sample terminal of the sample timer, and a second current-handling terminal, the sample gate providing a conductive path between the first and second current-handling terminals in response to the sample signal; a current-bias capacitance (C CB ) having a first terminal connected to the second current-handling terminal of the sample gate and a second terminal connected to a second reference voltage; and a voltage-to-current converter (34) having a control terminal connected to the first terminal of the current-bias capacitance, the voltage-to-current converter providing a ramp current proportional to a voltage level on the control terminal of the voltage-to-current converter. The converter of Claim 1, further comprising a second voltage-to-current converter (36) having a control terminal connected to the first terminal of the current-bias capacitance and having a current output terminal, the second voltage-to-current converter providing an output current on the output terminal, wherein the output current is proportional to a voltage level on the first terminal of the current-bias capacitance. The converter of Claim 1, wherein the output signal from the transconductance amplifier (GM1) has an amplitude proportional to the amplitude of the voltage level on the input terminal of the transconductance amplifier. The converter of Claim 1, wherein the input terminal of the transconductance amplifier (GM1) is one of first and second differential input terminals, and wherein the other of the first and second differential input terminals is connected to a fixed voltage. The converter of Claim 1, wherein the ramp capacitance circuit (20) further comprises: a current source (90) having a first terminal connected to the second terminal of the capacitance and a second terminal, the current source for alternately charging and discharging the capacitance; and a comparator circuit (70) having a first input terminal connected to a first reference voltage, a second input terminal connected to a second reference voltage, and an output terminal connected to the second terminal of the current source, the comparator circuit providing a first output on the output terminal when the voltage on the first terminal of the capacitance goes above the first reference voltage, thereby causing the current source to discharge the ramp capacitance, and a second output on the output terminal when the voltage on the first terminal of the ramp capacitance goes below the second reference voltage, thereby causing the current source to charge the ramp capacitance. The converter of Claim 5, further comprising a counter (80) having an input terminal connected to the comparator circuit, the counter providing a count representative of a number of times the capacitance is charged and discharged. The converter of Claim 6, wherein the counter (80) further comprises an output terminal, and wherein the counter is configured to provide an output signal on the output terminal indicating when the count is below a selected number. The converter of Claim 7, wherein the transconductance amplifier (GM1) further comprises an input-signal-override terminal connected to the output terminal of the counter, and wherein the transconductance amplifier is configured to provide a maximum output current when the output signal on the output terminal of the counter indicates that the count is below the selected number. A method of converting a frequency of a substantially periodic input signal to a reference current having a level proportional to the average period of the input signal, the method comprising: charging a ramp capacitance to a first reference voltage level in response to an edge of the input signal; changing the voltage level on the ramp capacitance using a current level that is a function of a current control signal indicating the frequency; and after the step of changing the voltage level on the ramp capacitance, comparing the voltage level on the ramp capacitance to a second reference voltage level and, based on the results of the comparing, adjusting the current control signal to adjust the current level used to change the voltage level on the ramp capacitance. The method of Claim 9, further comprising the step of converting the voltage on the reference capacitance to a current proportional to the voltage on the reference capacitance. The method of Claim 9, wherein the step of comparing includes holding the voltage level across the ramp capacitance constant during a sample period. The method of Claim 9, wherein the step of changing the voltage level on the ramp capacitance includes alternately charging and discharging the ramp capacitor. The method of Claim 12, wherein the first and second reference voltage levels are equal. A frequency-to-current converter, comprising a circuit (20) which is reset when an edge of the input frequency is received, said circuit comprising: a ramp capacitance having a first terminal and a second terminal; a current source (90) having first and second terminals, the first terminal connected to the first terminal of the ramp capacitance, the current source for alternately charging and discharging the capacitance, the current being determined by the frequency; and a comparator circuit (70) including: a first input terminal (V MAX ) connected to a first reference voltage; a second input terminal (V MIN ) connected to a second reference voltage; a third input terminal (V RAMP ) connected to the first terminal of the capacitance; and an output terminal connected to the second terminal of the current source; wherein the comparator circuit (70) provides a first output on the output terminal when the voltage on the first terminal of the capacitance goes above the first reference voltage, thereby causing the current source to discharge the capacitance, and a second output on the output terminal when the voltage on the first terminal of the capacitance goes below the second reference voltage, thereby causing the current source to charge the capacitance, the frequency determined current being adjusted according to the number of times the ramp capacitance is charged or discharged. The circuit of Claim 14, further comprising a counter (80) having an input terminal connected to the comparator circuit (70), the counter providing a count representative of the number of times the capacitance is charged and discharged. A frequency-to-current converter comprising: an input terminal (F INPUT ) for receiving a substantially periodic input signal; a ramp capacitance (C RAMP ) having first and second terminals; means for providing a first reference voltage (V START ) across the first and second terminals of the ramp capacitance; current means connected to the first terminal of the ramp capacitance, the current means for providing current (I RAMP ) to change the voltage across the first and second terminals of the ramp capacitance, wherein the current is controlled by a current control signal which indicates the frequency; and means for adjusting the current control signal in response to the voltage level on the ramp capacitance after a time period proportional to the period of the input signal. A frequency-to-current converter comprising: an input node (F INPUT ) for receiving a substantially periodic input signal; a ramp capacitance (C RAMP ) having first and second terminals; a reset circuit connected to the ramp capacitance, the reset circuit for periodically charging the ramp capacitance to a first reference voltage level; an output current generator (32, 90) connected to one of the first and second terminals of the ramp capacitance, the output current generator for establishing a discharge current level for the ramp capacitance, wherein the discharge current level is controlled by a discharge-current control signal which indicates the frequency; and differential circuit (GM1, 72, 74) connected to the ramp capacitance and to a second reference voltage level, the differential circuit for adjusting the discharge-current control signal in response to the voltage level on the ramp capacitance after a time period proportional to the period of the input signal.

Inventors: PHAM GIAO M, PHAM, GIAO M.
Applicants: SILICONIX INC, SILICONIX INCORPORATED
Application year: 1997
Main classification: H03K9
IPC codes: H03K9