A MULTI RESONANT RADIO ANTENNA

Abstract: A multi mode radio antenna is disclosed, operable in a multi-mode radio transceiver. One aspect of the present invention, provides a radio antenna having resonant frequencies operable to receive and transmit radio signals in different frequency bands according to two operating protocols. In accordance with another aspect of the invention, there is provided a method of operating the antenna.

Description: A MULTI RESONANT RADIO ANTENNA Field of Invention The present invention relates to a multi mode radio antenna and, in particular, relates to the same for use in a multi-mode mobile radio handset. Background Art Personal communication networks are being deployed extensively world-wide using cellular mobile radio systems. Earlier networks, still in operation, use analogue modulation formats for the radio air interface protocol. These analogue networks exhibit the problem of call saturation in high usage areas. To overcome this problem higher capacity air interface protocols using digital modulation format networks have been introduced in tandem, that is an area is covered by both systems. In the United States and Canada the early standardised analogue network known as AMPS has reached a fairly universal coverage of the populated North American continent. The newer digital networks, however, tend to be deployed in areas of high usage. A result of this is that there are areas of digital network coverage overlaying a universal analogue network coverage. Additionally, different air interface protocol standards of digital networks have been deployed regionally, since different telecommunications operators have developed their own protocols or have developed such protocols in line with national and sometimes international standards authorities, for example, the GSM protocol. Whilst it is reasonable to suppose that handsets operable for different radio communications protocols are similar from the users point of view, it is not possible, in particular, to use a digital mobile radio in an analogue cellular region and vice versa. This stems from the fact that whilst both types of handsets possess antennas, radio front end transmitter, receiver and baseband circuits, they operate on different air interface protocols which operate, inter alia at different radio carrier frequencies. Therefore it can be seen that each individual personal communications system user will need a dual network service for complete coverage. Consequently the user requires a handset that will not only function throughout the coverage area of the specific subscribed-to digital network, but also have a switched alternative mode to operate on the universal analogue network. The problem of implementing a dual mode handset has been considered to be surmountable by several different approaches; one solution uses two separate radio transceivers piggybacked and combined at the man-machine interface (keyboard and audio); a second solution uses two separate radio sections piggybacked and combined at the digital signal processing part of the radio transceiver, - applicants have a pending application relating to such a scheme, GB9603316.2. These two above approaches have problems in that both modes of operation transmit via an antenna. If the frequencies of operation are different, as indeed they will need to be, then two types of antenna will be necessary. For a dual mode terminal perhaps the simplest option for the antenna is to use a separate element for each of the desired bands. This could be in form of one external and one internal antenna, or two internal antennas. Two external antennas would be cumbersome, and unsightly. Such use of two antennas which have separate resonant frequencies of operation is accordingly complicated and unwieldy. One type of antenna is the monopole antenna which has a length corresponding to a quarter wavelength of its design frequency. This design is efficient, robust, provides a good bandwidth and, typically, can be a good match to a 50Ω input impedance. Figure 1 shows an example of such an antenna. Nevertheless the quarter wavelength monopole is relatively long and is limited to multiple frequency usage at the third and fifth harmonic frequencies. Whilst it may be possible to operate the antenna in two frequency bands associated with different radio systems, where the operating frequency of one band is a harmonic of the other operating frequency band, such an overlap would not be tenable because of the inevitable interference effects. Further, radio frequency spectrum allocation is not, typically, based upon the harmonics of a primary band. A number of dual band helical structures have been investigated at the Helsinki University of Technology, and these were presented at the 1996 IEEE VTC Conference. The helical structures presented are shown in figure 1. They consist of :- (a) two helical antennas, one within the other; (b) a helical-monopole combination; and (c) a helical antenna combined with a wound monopole. The paper states that the dual frequency operation can be obtained from all three of the- structures that are shown. Results for structure (a) state that it was tuned to the frequencies 1740 MHz and 900 MHz, and that 10 dB return loss bandwidths were obtained of 5.2% and 2.2% respectively. The dimensions for the antennas were Di = 6 mm, D2 = 3mm, D = 5 mm, Ih1 = 12 mm, I 2 = 14 mm, lm = 39 mm, l n = 13 mm, N = 5, N2 = 5, N3 = 7, and Is = 10 mm. Results for structure (b) state that it was tuned to the frequencies 1750 MHz and 894 MHz, and that 10 dB return loss bandwidths were obtained of 12% and 4.5% respectively. Structure (c) is simply a more compact version of (b), and not surprisingly has a narrower bandwidth. For the upper and lower bands, measured bandwidths of 11% and 2.9% were obtained where the overall structure height was 34 mm. Thus, in summary these antennas provide a bandwidth which is not sufficient for many radio applications, and also does not leave any margin for manufacturing tolerances. A dual band external antenna is described by Ali et al in 'A wide band dual meander sleeve antenna 1 , IEEE Antennas and Propagation Society International Symposium, 1995, vol.2 p.1 124-7, 18-23 June 1995, Newport Beach, CA, USA, and this is called the wide band dual meander sleeve antenna. This antenna is described as potentially useful as a low profile antenna for a dual mode handset. However, the results presented in the paper are for the case where the experimental antenna is mounted on a large ground plane (90 cm 2 ) and as such would not be suitable for applications such as mobile telecommunications handsets. Object of the Invention The present invention seeks to provide a multi-mode antenna which has a number of resonance bands and overcomes the aforementioned problems. The present invention also seeks to provide an antenna- arrangement for a multi-mode radio communications handset. The present invention further seeks to provide an antenna for a cellular radio transceiver which is aesthetically pleasing, low cost, of high strength and electrically efficient. Statement of the Invention In accordance with one aspect of the present invention, there is provided a radio antenna having resonant frequencies operable to receive and transmit radio signals in different frequency bands according to two operating protocols. In accordance with a second aspect of the invention, there is provided a dual resonance radio antenna operable at two frequencies, the antenna comprising a tubular element and a central element; wherein the tubular element is electrically connected to a feed element by way of a conductive base portion and has a rim defining a distal portion of the tubular element; wherein the central element extends from the base portion, within the tubular element and outwardly from the rim of the tubular element; wherein the electrical length from the feed point to the rim corresponds to a quarter of the wavelength of the higher operating frequency ; and wherein the electrical length from the feed point to the distal end of the central element corresponds to a quarter of the wavelength at the lower operating frequency. This design provides two operational frequency bands of wide bandwidth, especially at the higher frequency. Independent tuning of the respective wavebands is possible. It is to be noted that the reference to frequency band and frequency are used interchangeably within the specification for reasons of convenience, since the upper and lower resonant frequencies do not exist at spot frequencies, but rather across a range or band of frequencies. The central element is preferably coaxial. The base of the tubular element can be orthogonal with respect to the central element. The coaxial region of the tubular element can be filled or partially filled with a dielectric material. This can improve the robustness of the design. The distal portion of the central element can be encased within a dielectric material. The distal portion of the central element can be retractable within a proximal portion of the central element, coiled or be constructed of a flexible material. This makes the unit more compact or more easily stored. The longer wavelength section may not be employed if communication takes place at the higher frequency. The conductor element between the feed point and the base of the tubular element can be extended whereby the capacitive effect of the base of the tubular element with respect to the ground plane is reduced, which changes the an input impedance. This feature can assist in the matching of the antenna with respect to a feedline of a given input impedance. The tubular element can comprise a conductive braid and/or can be cylindrical. By increasing the diameter of the coaxial section, the bandwidth at the higher frequencies could be increased, but this would reduce the efficiency at the lower frequencies of operation. In accordance with another aspect of the invention, there is provided a dual resonance radio antenna operable at two wavelengths λ - and λ2, where λ -\ corresponds to the higher frequency, comprising a first monopole element having a length corresponding to a quarter of the wavelength of the lower operating frequency and a cylindrical element comprising a coaxial stub which surrounds the first monopole element having an electrical length corresponding to λ -\/4 the cylindrical portion being electrically connected at the top of the monopole, providing the monopole with an exposed λ 1/4 length between the stub and a base of the antenna; wherein at the lower operating frequency, current is induced on the outer surface of the coaxial stub which is in phase with the exposed λ -\l4 section and the antenna performs as a quarter wave monopole; and wherein at the higher operating frequency the stub presents a high impedance at the top of the monopole whereby the effective length of the monopole is λ ι/4, as measured from the base of the antenna. Advantageously, the coaxial region of the stub is partially or completely filled with dielectric. In accordance with a fourth aspect of the invention there is provided a multi-resonant antenna comprising first and second coupled lines operable to transmit and receive radio signals, a feed and a ground plane; wherein the feed provides radio signals to the first line, which line extends relative to the ground plane, and; wherein the lines are coupled such that at a first frequency, the phase velocity of the surface currents across the coupled lines are increased and, at a second, higher frequency, the phase velocity of the surface currents across the coupled lines are decreased. The coupled transmission lines can be coupled at a distal end of the first transmission line (conductive element); said second line can be parallel with said first line. A third coupled line can be present, which third line can be parallel with said first line. In accordance with a fifth aspect of the invention there is a provided a multi-resonant antenna comprising first, second and third coupled lines operable to transmit and receive radio signals, a feed and a ground plane; wherein the feed provides radio signals to the first line, which line extends relative to the ground plane, and; wherein the lines are coupled such that at a first frequency, the phase velocity of the surface currents across the coupled lines are increased and, at a second, higher frequency, the phase velocity of the surface currents across the coupled lines are decreased. A fourth coupled line may be provided, which fourth line can be parallel with said first line. A fifth coupled line may be provided, which fifth line can be parallel with said first line. In accordance with a sixth aspect of the invention there is a provided a multi-resonant antenna comprising adjacently placed conductive lines, which lines have a Schiffman phase frequency response whereby, at a high frequency mode of operation, the phase velocity of surface currents is reduced and at a lower frequency mode of operation, the phase velocity of surface currents is increased. In accordance with a further aspect of the invention, there is provided a method of operating a dual resonance radio antenna operable at two wavelengths λ -\ and A 2, where λ -\ is the higher frequency, the antenna comprising a tubular element and a central element; wherein the tubular element is electrically connected to a feed element by way of a conductive base portion and has a rim defining a distal portion of the tubular element; wherein the central element extends from the base portion, within the tubular element and outwardly from the rim of the tubular element; wherein the electrical length from the feed point to the rim corresponds to a quarter of the wavelength at the higher operating frequency ; and wherein the electrical length from the feed point to the distal end of the central element corresponds to a quarter of the wavelength at the lower operating frequency; wherein, in a receive mode, at the higher frequency, the antenna portion from the feed point to the distal rim of the tube portion of the antenna resonates as a quarter wave monopole and the coaxial region between the tube and the central element appears as a quarter wavelength short circuited stub, thereby preventing the excitation of currents along the remaining section of the central element and whereby the antenna operates in a fundamental mode; and at the lower frequency, the antenna resonates in a fundamental mode of operation over the whole of the antenna length; the overall length is approximately λ2 and so a second resonance is generated as surface currents on the outer surface of the tube induce currents on the central element protruding from the tube and which are in phase with the currents on the outer surface of the tube, whereby the structure resonates as a quarter wavelength monopole at the lower frequency, and, in each case the signals are received by a receiver apparatus by way of the feed point. In accordance with a still further aspect of the invention, there is provided a method of operating a dual resonance radio antenna operable at two wavelengths λ i and A.2, where λ -\ is the higher frequency, the antenna comprising a tubular element and a central element; wherein the tubular element is electrically connected to a feed element by way of a conductive base portion and has a rim defining a distal portion of the tubular element; wherein the central element extends from the base portion, within the tubular element and outwardly from the rim of the tubular element; wherein the electrical length from the feed point to the rim corresponds to a quarter of the wavelength at the higher operating frequency ; and wherein the electrical length from the feed point to the distal end of the central element corresponds to a quarter of the wavelength at the lower operating frequency; wherein, in a transmit mode, at the higher frequency, the antenna portion from the feed point to the distal rim of the tube portion of the antenna resonates as a quarter wave monopole and the coaxial region between the tube and the central element appears as a quarter wavelength short circuited stub, thereby preventing the excitation of currents along the remaining section of the central element and whereby the antenna operates in a fundamental mode; and at the lower frequency, the antenna resonates in a fundamental mode of operation over the whole of the antenna length; the overall length is approximately A.2 and so a second resonance is generated as surface currents on the outer surface of the tube induce currents on the central element protruding from the tube and which are in phase with the currents on the outer surface of the tube, whereby the structure resonates as a quarter wavelength monopole at the lower frequency, and, in each case the signals are transmitted by a transmitter apparatus by way of the feed point. In accordance with a still further aspect of the invention, there is provided a dual mode radio transceiver operable on either of two frequency bands, including an antenna arrangement for transmission and reception and comprising a tubular element and a central element; wherein the tubular element is electrically connected to a feed element by way of a conductive base portion and has a rim defining a distal portion of the tubular element; wherein the central element extends from the base portion, within the tubular element and outwardly from the rim of the tubular element; wherein, in use, at the higher frequency, the tubular element interacts with the central element to provide a short circuit whereby the antenna resonates between the rim of the tubular element and the feed; and at the lower frequency, the antenna operates in a fundamental mode of operation over the whole of the antenna length. In accordance with a further aspect of the invention, there is provided a method of operating a dual resonance radio antenna operable at two wavelengths λ ι and A 2, where λ -\ corresponds to the higher frequency, the antenna comprising a first monopole element having a length corresponding to a quarter of the wavelength of the lower operating frequency and a cylindrical element comprising a coaxial stub which surrounds the first monopole element having an electrical length corresponding to λ 1/4 the cylindrical portion being electrically connected at the top of the monopole, providing the monopole with an exposed λ 1/4 length between the stub and a base of the antenna; wherein at the lower operating frequency, current is induced on the outer surface of the coaxial stub which is in phase with the exposed A ι/4 section and the antenna performs as a quarter wave monopole; and wherein at the higher operating frequency the stub presents a high impedance at the top of the monopole whereby the effective length of the monopole is A /4, as measured from the base of the antenna. In accordance with another aspect of the invention there is provided method of operating a multi-resonant antenna, said multi-resonant antenna comprising first and second coupled lines operable to transmit radio signals, a feed and a ground plane, wherein the first line extends relative to the ground plane; wherein, in a transmit mode, the method comprises the steps of providing radio signals, via the feed, to the first line, wherein the lines are coupled such that at a first frequency, the phase velocity of the surface currents across the coupled lines are increased and, at a second, higher frequency, the phase velocity of the surface currents across the coupled lines are decreased; whereby the lines resonate and the signals are transmitted via the lines. In accordance with a still further aspect of the invention there is provided a method of operating a multi-resonant antenna, said multi- resonant antenna comprising first and second coupled lines operable to receive radio signals, a feed and a ground plane, wherein the first line extends relative to the ground plane; wherein, in a receive mode, the method comprises the steps of receiving radio signals, via the coupled lines such that at a first frequency, the phase velocity of the surface currents across the coupled lines are increased and, at a second, higher frequency, the phase velocity of the surface currents across the coupled lines are decreased; whereby the lines resonate and the coupled radio signals are output via the feed. Brief Description of Drawings In order that the present invention can be more fully understood and to show how the same may be carried into effect, reference shall now be made, by way of example only, to the Figures as shown in the accompanying drawing sheets wherein: Figures 1 a 1 c depict three dual frequency antenna configurations; Figures 1 d 1 g depict two antenna configurations and radiation patterns therefor; Figure 2 depicts a typical handset schematic; Figure 3 is a detailed implementation of a dual mode radio front end; Figures 4 and 5 show a first dual band antenna structure made in accordance with the invention; Figure 6 shows the measured return loss for the antenna shown in Figures 4 and 5; Figure 7 shows the measured azimuth radiation pattern for the dual band monopole shown in Figure 5 at 860 MHz; Figure 8 shows the measured azimuth radiation pattern for the dual band monopole shown in Figure 5 at 1920 MHz; Figure 9 shows the measured elevation radiation pattern for the dual band monopole shown in Figure 5 at 860 MHz; Figure 10 shows the measured radiation pattern for the dual band monopole shown in Figure 5 at 1920 MHz; Figure 1 1 shows a second embodiment of an antenna made in accordance with the invention; Figure 12 shows a third embodiment of an antenna made in accordance with the invention; Figure 13 is a schematic diagram of a fourth dual band monopole structure made in accordance with the invention; Figure 14 is a first embodiment of an antenna made in accordance with the invention; Figure 15 shows the measured return loss for the dual frequency monopole prototype shown in Figure 14; Figure 16 shows the measured azimuth radiation pattern for the dual band monopole shown in Figure 14 at 900 MHz; Figure 17 shows the measured azimuth radiation pattern for the dual band monopole shown in Figure 14 at 1900 MHz; Figure 18 is a schematic diagram of a fifth embodiment of an antenna made in accordance with the invention; Figure 19 shows the dimensions of a fifth embodiment of an antenna made in accordance with the invention; Figure 20 shows the measured return loss for the dual frequency monopole prototype shown in Figure 19; Figure 21 shows the measured azimuth radiation pattern for the dual band monopole shown in Figure 19 at 900 MHz; Figure 22 shows the measured azimuth radiation pattern for the dual band monopole shown in Figure 19 at 1900 MHz; Figure 23 shows a sixth embodiment of the invention; Figure 24 shows a seventh embodiment of the invention; Figure 25 shows an eighth embodiment of the invention; Figure 26 shows a ninth embodiment of the invention; Figure 27 shows a tenth embodiment of the invention; Figures 28 & 29 show various embodiments of the invention; Figure 30 shows approximate dimensions for the lengths of the antenna elements of the fifth embodiment; Figure 31 a - j show graphical performance data; Figures 32 (i) shows an exemplary Schiffman phase shifter phase response as a function of frequency for a conductive C-section, 15 (ii); Figures 33 a-e show a co-ordinate system and gain and cross- polarization levels relating to the tenth embodiment at two frequencies; Figure 34 shows a further embodiment; Figure 35 shows a Smith chart for the embodiment of Figure 34; Figure 36 shows the S1 1 plot of the embodiment of Figure 34; Figure 37 shows a still further embodiment; Figure 38 shows the S11 plot of the embodiment of Figure 37; Figure 39 shows a Smith chart for the embodiment of Figure 37. Detailed description of the preferred embodiments There will now be described by way of example the best mode contemplated by the inventors for carrying out the invention. In the following description, numerous specific details are set out in order to provide a complete understanding of the present invention. It will be apparent, however, to those skilled in the art that the present invention may be put into practice with variations of the specific. Widespread use has been made of the quarter wavelength monopole antenna, which will radiate with respect to a ground plane. In the case of mobile communications handset antennas, the ground plane is provided by the casing associated with the handset electronics enclosure. For analysis purposes, the quarter wavelength monopole, such as is depicted in Figure 1 d may be viewed as possessing a quarter wavelength image and it forms the half wavelength equivalent, as shown in Figure 1 e. Figure 1f shows the resulting current distribution pattern along the length of a linear monopole antenna, for several wavelengths corresponding to λ/4 and 3λ/4, where λ is the free space wavelength. Figure 2 shows a block diagram of a typical cellular radio handset. Radio frequency signals are received and transmitted by the antenna 2 which is connected to a radio front end 4. In the radio front end transmit and receive signals are converted between radio frequency and base band, whereby digital signal processing means 6 encode the transmit and decode the receive signals and from these can determine the audio signals which are communicated to and from the handset user by loudspeaker 7 and microphone 8. The front end will typically contain transmit and receive paths which are mixed to an intermediate frequency with a local oscillator. These intermediate frequency signals will be further processed and mixed so that the input and output signals to and from the front end are at baseband and suitable for digital to analogue or analogue to digital conversion, as appropriate, prior to digital signal processing. Referring now to Figure 3, there is shown a handset architecture, comprising a dual mode radio front end for the reception of both digital PCS 1900 signals and analogue AMPS signals. PCS 1900 operates in the frequency band 1930 to 1990 MHz on the receive downlink to the handset and in the 1850 to 1910 MHz band on the transmit uplink from the handset. AMPS operates in the frequency band 824 to 849 on the transmit uplink from the handset and in the 869 to 894 MHz band on the receive downlink to the handset. PCS 1900 operates either in an uplink mode or in a downlink mode; AMPS can operate in both modes simultaneously. For this reason the switch 14 from the antenna 12 has three positions. Details of the antenna are not shown in this figure for simplicity. Turning now to the receive path for the digital PCS 1900 signals, when the switch 14 directs incoming digital PCS 1900 signals to the PCS 1900 receive path, the signals from the band select filter 22 are passed to a mixer 30 which mixes the received signal with a signal from a synthesised local oscillator 34 to produce an intermediate frequency (IF) signal at 225 MHz which is subsequently amplified by further amplifying means 36. The PCS 1900 signals are passed through a second switching circuit 44 which operates simultaneously with the first switch 14 by mode control means (not shown). The mode control means identifies whether the signals are digital or analogue modulation and determines in which mode the transceiver is operating. The receive signal output from which 44 is fed to an IF amplifier with automatic gain control and a receive signal strength indicator (RSSI). If an analogue AMPS radio signal were present at the antenna and a decision made to receive that signal, the switch 14 would feed the signal from the antenna 12. For transmit, the PCS 1900 and AMPS baseband signals are raised to 150 MHz and 225 MHz intermediate frequencies (IFs) respectively. The upconverted IF containing either the PCS 1900 signal at 150 MHz or the AMPS signal at 225 MHz is applied respectively to the PCS 1900 transmit band at 1850 to 1910 MHz and the AMPS transmit band at 824 to 849 MHz. The respective signals are RF band filtered by 26 and 28 prior to power amplification and then fed to the antenna via separate filters and switch 14. The main factors that should be taken into account in the design of an antenna are electrical performance, volume required (internally), cost, and manufacturability. With regard to the electrical performance of antennas, the main performance parameters are:- radiation efficiency; isolation (where two elements are used); typically the return loss should be >10 dB across the operating band. Thus the PCS antenna requires a 7.3% 10 dB return loss bandwidth, while the AMPS antenna requires a 8.1% bandwidth. Mean effective gain is a measure of the handset antenna radiation pattern, and involves the multi-path angular density function. SAR is fixed by regulatory limits. Radiation efficiency, this should be greater than -2 dB for the handset in isolation (ideally >- 1 dB for external antennas). With the handset in the presence of the head and hand the efficiency should be >-3 dB. The isolation required between two antenna elements ought to be >10 dB, since if the coupling is too high this can result in a significant reduction in efficiency. Referring now to Figures 4 and 5, a schematic diagram of a dual band monopole structure made in accordance with the invention is shown. The antenna 116 comprises a conductive tube 118 having a distal rim 120, an insulator element 122 and a central axial element 124, which insulation extends beyond the rim 120 of the tube 118. The tube and coaxial element can conveniently be manufactured from brass or similar metals; PTFE can conveniently be employed as an insulator since it is of low weight and low relative permittivity. The wavelengths for the two resonances are given by λ -\ and A 2, where A 1 corresponds to the higher frequency. For tests, the antenna was mounted upon a conductive box 128 which simulated the ground effects similar to those provided by a shielding plane surrounding the electronics of a mobile radio handset. A transceiver was connected to an sma connector 130 mounted on a ground portion 128. The sma connector exchanging signals with the antenna via the fed point There is no d.c. connection between the antenna 1 16 and the case 128. With reference to Figure 3, appropriate matching circuits could be placed between the antenna switch and the filters 16, 18 and 20, but each particular arrangement can be set up in any one of several ways as will be known to those skilled in the art. Figure 5 shows the dimensions in mm of the first embodiment. At the higher frequency, the distance from the feed point to the distal rim of the tube portion of the antenna simply appears as a quarter wave monopole. The coaxial region between the tube and the central element appears as a quarter wavelength short circuited stub; the coaxial stub creates an open circuit at the rim 120 which chokes the currents, thereby preventing the excitation of currents along the remaining section of the central element and whereby the antenna operates in a fundamental mode. At the lower frequency, the antenna operates in a fundamental mode of operation over the whole of the antenna length. The overall length is approximately A 2 4 and so a second resonance is generated. Surface currents flowing on the outer surface of the tube are now no longer choked since the coaxial stub is now approximately A 2 /8 long. This is equivalent to an inductive reactance at the rim of the tube which affects the input impedance. Currents are now induced on the central element protruding from the tube and these will be in phase with the currents on the outer surface of the tube. Therefore the structure radiates like a quarter wavelength monopole at the lower frequency. The measured return loss is shown in Figure 6. The return loss plot indicates that, in the higher frequency range, this single element provides a suitable bandwidth for many present systems, such as the PCS1900 standard (1850 - 1990 MHz), where a 10 dB level across this band is achieved. In fact, the 10 dB level extends to 2500 MHz, making the antenna suitable for many applications such as DCS1800 (1710 - 1880 MHz) and the proposed third generation standards such as the Universal Mobile Telephone System (UMTS) as proposed by ETSI or the Future Public Land Mobile Telecommunications System (FPLMTS) as proposed by the ITU which have both been assigned the 1885 - 2200 MHz radio frequency spectrum. For the lower resonance, there is a 7.5 dB return loss bandwidth from 820 - 1130 MHz. This provides an adequate bandwidth for AMPS the frequency range of which extends from 824 MHz - 894 MHz, whilst a 10 dB bandwidth is provided for the GSM range, the frequency range of which extends from 890 to 960 MHz (a reduction in the band at around 920 MHz corresponds with a guard band of 915 - 925 MHz and is therefore inconsequential). Nevertheless, retuning can easily be performed, as those skilled in the art would be familiar, by adjustment of the length of the central conductive element. The measured radiation patterns for azimuth and elevation, at 860 MHz and 1920 MHz are shown in Figures 7, 8, 9 and 10. The azimuth radiation patterns are omnidirectional as is the case for a A/4 monopole antenna. This design does not rely upon there being a frequency ratio of two as is necessary for some dual resonance designs. One aspect of the design which does not improve the performance at the lower operating frequency is the stub which provides an inductive reactance at the open end at the lower frequency. This can affect the input impedance such that some matching is required. There are two ways of overcoming this: either a matching element can be positioned in the feed circuitry, which can be inconvenient, or the stub can be fully or partially dielectrically loaded which changes the reactive component in the coaxial region whereby the electrical length of the inside section may be maintained for a shorter physical length, bearing in mind that the closed end of the tube must be closer to the feed than the rim. It is the distance from the feed point along the outside of the tube to the distal rim 120 of the tube 1 18 which determines the electrical length of the antenna, which corresponds to a quarter of a wavelength at the higher frequency. Therefore, if the dielectric constant of the dielectric insert within the tube is high, then the tube can be shorter since there must be an electrical length of A 1 14 between the electrical short created at the bottom of the tube and the rim whereby an effective choke is present at the higher frequency. If the physical length is much shorter than the electrical length within the tube, then the tube section must be positioned a distance from the feed point since the electrical length along the outside surface of the tube and central element is equivalent to the physical length, in terms of wavelength. This is beneficial with respect to the capacitive effect induced by the closed end of the tube with respect to ground, which is reduced with respect to the design shown in Figures 4 and 5, whereby the antenna structure may more easily be matched. The dielectric loading within the tube does not affect the overall length of the antenna; only the position of the tube section relative to the central element, as shown in Figure 11. A further advantage obtained by employing a dielectric insert is that there is an increase in the mechanical rigidity of the tube. Figure 12 shows an antenna which has continuous dielectric loading 124 within the tube 1 18. The dielectric insert must be of an appropriate dielectric constant whereby the electrical length inside the tube corresponds to a quarter of the wavelength at the higher frequencies. Thus, with respect to Figures 1 1 and 12, the dielectric constant of the shorter tube, in Figure 11 has a higher dielectric constant whereby the electrical length within the tubes is the same for both antennas. The distal portion of the central element 122 can also be enclosed by a dielectric whereby the physically sharp end may be encased to provide convenience for users of mobile communication handsets, whilst simultaneously providing an improved aesthetic qualities of the design. Furthermore, the electrical length at the lower frequency is reduced, whereby the overall length of the antenna is reduced. For convenience the central element could be made such that it is flexible to a certain extent. Antennas comprising tightly twisted/coiled wire, as are known are particularly suitable. If it was desired to reduce the length of the antenna, it would be possible to coil the distal section of the central section whereby the distal section is reduced in length. This would result in a reduction in the bandwidth available, but this is possible for certain scenarios. An alternative option would be to have a slideably retractable central element portion, whereby for use at the higher frequencies and for storage the antenna is of reduced length and for use at lower frequencies, the antenna may be extended. It is possible to place a material with a similarly high dielectric constant around the central element from the feed to the base of the tube, to reduce the physical length yet retain the electrical length. Nevertheless, a reduction in bandwidth at the higher frequency may result from placement of such additional dielectric material. It should be remembered that because the tube base portion as seen in the Figures 4,5 1 1 and 12 is orthogonal with respect to the central element, that the physical path and hence the electrical path on the outside of the tube are greater than the distance of the base of the tube to the rim of the tube (which affects the internal electrical length). Of course, it would be possible to have a rounded or angled base to the tube to ameliorate such effects. Referring now to Figure 13, a schematic diagram of a dual band monopole structure made in accordance with the invention is shown. The wavelengths for the two resonances are given by A i and A 2, where A i corresponds to the higher frequency. At the higher frequency the antenna simply looks like a quarter wave monopole. This is because there is a A 1/4 coaxial stub at the top of the initial A 1/4 'monopole'. The stub is short-circuited at one end, presenting an open circuit at the top of the monopole. For the lower frequency, current is induced on the outer surface of the coaxial stub which is in phase with the lower A i/4 section. The overall height is A 2 4 and so a second resonance is generated. By the use of a dielectric loading, the second section of the antenna can be varied in length. The dimensions of one antenna are shown in figure 14. The antenna was mounted on a rectangular PCB with dimensions comparable to a standard handset. The measured return loss is shown in figure 15. For the lower resonance the 10 dB return loss bandwidth is 800 - 930 MHz. The centre of this band is 865 MHz, and using this centre frequency the percentage bandwidth is 15%. Note that the AMPS band (824 - 894 MHz) is accommodated within this bandwidth. For the upper resonance the 10 dB return loss bandwidth is 1870 - 2050 MHz. The centre of this band is 1960 MHz, and using this centre frequency the percentage bandwidth is 9.2%. While this bandwidth is adequate for the PCS 1900 band (1850 - 1990 MHz), some slight retuning is required, which would involve a small lengthening of the 33 mm monopole section shown in figure 15. The measured azimuth radiation patterns at 900 MHz and 1900 MHz are shown in figures 16 and 17. The measured azimuth gain is quite low at 900 MHz, and further measurements are required to determine the antenna efficiency, and elevation patterns in the two bands. The elevation pattern for 900 MHz may well be down tilted. At the lower frequency, the overall structural length is designed to be nominally equivalent to a quarter of a wavelength long. This is where the reason for the choice of the frequency ratio of two becomes apparent: it is a quarter of the wavelength at the higher frequency. Unfortunately, the stub cannot be ignored. This is now an eighth wavelength short circuited stub which results in an inductive reactance at the open end. Consequently, the structure looks like a quarter wavelength monopole with an inductive reactance at its centre. This affects the input impedance such that some matching is required. There are two ways of overcoming this: either a match can be placed on the board which is inconvenient, or the stub can be partially dielectrically loaded which changes the reactive component in the coaxial region but does not affect the outside whereby the structure can fairly easily empirically be matched. Figure 18 shows a second embodiment 90 with a 1 mm thick tube 92, closed at a distal end and having a PTFE plug 94 inserted at the open end, a copper tube 96 extending from a sma connector 98 mounted on a ground portion, such as a mobile phone case, 100. There is no d.c. connection between the tube 96 and the case 100. The tube could be replaced with a solid element: alternatively the antenna structure could be made flexible. Figure 19 shows the physical dimensions of an embodiment. The flange of the dielectric protrudes by 2 mm in the example shown. This enables the structure to be self-locating since it ensures that the dielectric extends to a particular depth inside the structure. Nevertheless, it could be flush. The coaxial region can also be fully dielectrically loaded but the electrical length would be changed, which would require a shortening of the coaxial region, but it shortens the whole structure. It does nothing to change the length of the intermediate portion operable at the highest frequency because the open circuit still exists. By increasing the lowest frequency some control over the frequency ratio can be obtained. This allows a consequential flexibility in frequency ranges so that frequency combinations such as GSM and DECT, AMPS and PCS 1900, can be covered. Equivalent performance is obtained in the azimuth pattern at both frequencies . The measured return loss is shown in Figure 20; it can be seen that the return loss for this element is greater than 10 dB across both the AMPS band and the PCS bands. The square markers show the band limits. Figure 21 shows predicted and measured vertical polarisation patterns for a quarter wave monopole in the same position as the dual resonance one and for the dual resonance monopole at 860 MHz. The results show that an antenna made in accordance with the invention provides equivalent performance to a quarter wavelength monopole. The azimuth pattern for the dual resonance antenna is vertically polarised and omnidirectional, with a mean gain of 0 dBi and a pattern ripple of 3.3 dB and you can see they are very close, even in the elevation plane. Figure 22 shows the corresponding results at the higher frequency, 1920 MHz, and the difference between the predicted and measured elevation patterns is almost non-existent. If it was desired to reduce the length of the monopole, it would be possible to coil the lower section up whereby the lower section is reduced in height. This would result in a reduction in the bandwidth available, but this is possible for certain scenarios. A suitably flexible coaxial stub could also be employed to coil the top section as well but that would have to be quite wide. These embodiments of the present invention thus provide an antenna design which has good bandwidth for several frequency bands; the antenna can be manufactured such that the overall dimensions can be reduced; the antenna can be provided with a dielectric surround whereby the antenna can be made sufficiently rugged whereby it is not susceptible to damage in normal use, the dielectric surround and the conductive elements preferably being suitably elastic and flexible. The sixth embodiment, shown in Figure 23, is a two dimensional equivalent of the antenna depicted in Figure 14. The printed antenna comprises a feed part 602 from which a first elongate printed member 604 extends. Second and third elongate members 606 extend parallel on either side of the first member, these second and third members being fed by a member 608 perpendicularly attached to a distal end of the first member. The feed part 602 lies adjacent a ground plane 610 associated with, for example a handset enclosure, and would be connected to a radio frequency transceiver. Figure 24 is a seventh embodiment of the invention and differs from the fifth embodiment in that two second and third arms 706 are not parallel but diverge from the distal end, and in that fourth and fifth arms 710 lie parallel with the first member 704, said fourth and fifth arms being attached to the first member by connecting members 712. Such divergence of the arms 706 from the distal end reduces coupling between the second and third arms and the fourth and fifth arms and was found to improve the impedance of the structure at higher frequencies. Figure 25 is an alternative to this design in that there are no third and fifth arms and that the second arm 806 is parallel with the first member 804. The ninth embodiment, as shown in Figure 26, is a still further variant of the design of Figure 24; second 906 and third 910 arms lie on the same side of the first element 904 whereby lateral dimensions are reduced. Figure 27 shows an antenna similar to the ninth embodiment (Figure 26) but has a stub element 1014 which was found to improve matching. Figures 28 - 29 show two other suitable configurations of antenna which can perform in a multi-resonant fashion which further variants can have triangular elements. Whilst being compact in the longitudinal dimensions of the first element from the feed point, the lateral dimensions are increased - which of course may be acceptable depending upon the overall design requirements of the dual band installation. Examples can be conveniently manufactured employing printed copper tracks on a dielectric substrate such as FR4. Flexible dielectric substrates can be employed which, in the case of a mobile communications handset, would enable the antenna to be flexible, which in turn could be more appealing to the end user. In order to reduce the antenna length (analogous to the height of the antenna when used as a handset antenna) it is possible to curve the structure but this increases the lateral dimensions with consequential changes being necessary for manufacturing requirements. Referring now to Figures 30 onwards, the operation of this type of embodiment of antenna will now be discussed. Figure 30 shows approximate dimensions for the lengths of the antenna elements of the tenth embodiment for operation at 824 - 894 MHz and 1.850 - 1.99 GHz frequency band of operation. This embodiment was tested within an anechoic chamber and was subject to numerical electromagnetic code moment method computer simulation using computer test and analysis programs known under the WIPL brand. Figure 31 a shows a Smith chart and Figure 31 b shows an S1 1 plot for this antenna in the frequency range 0.5 - 2.5 GHz. The performance in Figures 31 c - f show the real and imaginary current distributions, which have the form of the third harmonic, at 2.0 GHz for the antenna elements 1004, 1008, 1006 and 1010 respectively. When scaled relative to the actual lengths of the antenna it can be seen that the antenna structure provides a decreased phase velocity relative to free space. This lowers the resonance from that expected with respect to the structure shown in Figure 13. In contrast, and with reference to Figures 31 g - j, the real and imaginary current distributions at 0.9 GHz are shown. These have the form of the first harmonic or fundamental. When scaled relative to the actual lengths of the antenna, it can be seen that the antenna structure provides an increased phase velocity relative to free space. At two particular frequencies, 900 MHz and 1.85 GHz, it was calculated that the antenna structure had an apparent length of 0.33 λ and 0.678 λ, respectively. Typically these should be 0.25 λ and 0.75 λ for a straight monopole. This result can be explained with reference to a Schiffman phase frequency response. The Schiffman effect is observed in a coupled transmission line structure having two of its ends connected together, the input impedance Z ιn of which being determined by the equation: Z in = ( Z^ . Z 0o ) 1 2 , where Z 0e and Z 0o are the even and odd - mode characteristic impedances, respectively of the coupled section. Antennas made in accordance with this embodiment of the invention have non - uniform characteristic impedances along the coupled transmission lines formed between the first and second antenna members which are parallel or divergingly spaced apart from a coupled point (such as the distal end of the first antenna member). Since the antennas made in accordance with the invention extend perpendicularly from a ground plane: this means that the characteristic impedance of the antenna elements varies as the structure projects outwardly and thus can affect the coupling between the two transmission lines. Referring now to Figure 32 (i), there is shown a Schiffman graph (transmission phase - frequency response plot) for a homogeneous section shown in Figure 32 (ii): at 860 MHz, I = 120°, i.e. f3 on the graph is 60° and, assuming a typical ratio of Z 0e / Z 0o = 4, we obtain a transmission phase change of 80°; a phase delay of 86° would be sufficient to allow the monopole to resonate. The structure provides increased phase velocity relative to free space. At 1.85 GHz, I = 244°, τ on the graph is 122° and a transmission phase delay of approximately 285° is obtained, although a figure closer to 259° would be preferred. The structure provides a reduced phase velocity relative to free space at this frequency. The Schiffman effect can enable a variation of operable frequency bands for an antenna provided there is adequate matching. Thus the Z 0e / Z^ value is controllable by varying the spacing between the coupled transmission lines. The antennas that have been made have proved to be somewhat shorter than pure theory would suggest. Turning now to Figures 33, there are shown selected radiation patterns relating to the fifth embodiment at 2.0 GHz and 900 MHz. For reference purposes, Figure 33a explains the spherical co-ordinate system employed. Note that the scale on the graphs for the circular coordinates refer to ϋ = 180° for the vertically downward direction and r = 0°for the vertically upward direction. The gain and cross-polarization levels are shown in Figures 33b and 33c for the azimuth pattern and the elevation pattern at 2.0 GHz, respectively. The gain is omni- directional + 10%, (+1 dB); the cross-polarization levels are low, being of the order of 20 dB lower than the co-polar levels. Figures 33d and 33e show the gain and cross-polarization levels for the azimuth pattern and the elevation pattern at 900 MHz, with the gain again being omnidirectional + 10%, (+1 dB); and the cross-polarization levels being low. Referring now to Figure 34, there is shown a further embodiment of the invention which was fabricated on a 2.5 mm thick FR4 dielectric substrate 172 and has dimensions as detailed. The antenna 170 comprises a general unequal U-shape copper track, with a copper wire 174 of 0.84 mm diameter lying parallel to the arms 176, 178 of the U and spaced therebetween, being connected to the connecting section 180 of copper track between the arms. A matching element 182 comprising a small length copper wire is positioned proximate the feed point 184. The antenna was mounted on a 150 X 55 X 20 mm box (not shown) to simulate the ground characteristics of a radio telecommunications handset, the copper tracks being fed via an sma connector (not shown) coupled to a microwave test generator. Further tuning of the antenna is possible for the design shown, as is within the scope of a skilled person, whereby the output of the antenna can be tailored to a particular requirement. A Smith chart is shown in Figure 35 and an S1 1 plot is shown in Figure 36. Figure 37 shows a still further embodiment with dimensions as detailed, the antenna 200 comprising a general W-shape, with the central arm 202 being the longest and being connected to an sma connector feed 204 and the outside arms 206, 208 being of different length, being connected at 210 to the central conductor. As is the case with the antenna described above, a matching element 212 is positioned proximate the feed point, with a copper wire of 0.93 mm diameter connecting the matching element to the central conductor, the antenna being mounted on a box as above. An S1 1 plot is shown in Figure 38: four modes of resonance are apparent, at the following frequencies: 720 MHz, 870 MHz, 1520 MHz and 1890 MHz. At the lower frequency, mode of operation, the wavelength is longer than that of the higher frequency signals and there is a greater phase velocity. Accordingly, the resonances are closer together. Conversely, at the higher frequency mode of operation, the wavelength is shorter and the phase velocity is reduced relative to the lower frequency and two higher frequency resonances are further apart in terms of frequency spacing. A corresponding Smith chart is shown in Figure 39. Although the transmission lines of the embodiments have been made from plated dielectric substrates, the transmission line structures could comprise other types of conductive materials, such as metallic wires. An advantage, if the antenna is fitted to a mobile communications handset is that if the antenna is broken, then it is more likely to be easily and cheaply replaced. If made on a flexible dielectric support structure, then it would be less liable to fracture when carried. Alternatively, the dielectric supporting the antenna could be slideable relative to a handset casing.

Claims: CLAIMS 1. A multi resonant radio antenna having operable to receive and transmit radio signals in different frequency bands according to two operating protocols. 2. A multi resonant radio antenna according to claim 1 , comprising a tubular element and a central element; wherein the tubular element is electrically connected to the central element which extends from within the tubular element and outwardly from a rim of the tubular element; and the central element is electrically connected to a feed element. 3. A multi resonant radio antenna according to claim 2, wherein the electrical length from the feed point to the rim of the tubular element corresponds to a quarter of a wavelength at a first operating frequency ; and wherein the electrical length from the feed point to the distal end of the central element corresponds to a quarter of the wavelength at the lower operating frequency. 4. A multi resonant radio antenna according to claim 2 or 3, wherein the rim of the tubular element and the central element are operable to provide an open circuit at one frequency and a closed circuit at another frequency. 5. A multi resonant radio antenna according to any one of claims 2 - 4, wherein the tubular element is electrically connected to a feed element by way of a conductive base portion and has a rim defining a distal portion of the tubular element; wherein the central element extends from the base portion, within the tubular element and outwardly from the rim of the tubular element; wherein the electrical length from the feed point to the rim corresponds to a quarter of the wavelength of the higher operating frequency ; and wherein the electrical length from the feed point to the distal end of the central element corresponds to a quarter of the wavelength at the lower operating frequency. 6. A multi resonant radio antenna according to any one of claims 2 - 5 wherein the distal portion of the central element is retractable within a proximal portion of the central element. 7. A multi resonant radio antenna according to any one of claims 2 - 5 wherein the distal portion of the central element is coiled. 8. A multi resonant radio antenna according to any one of claims 2 - 5 wherein the distal portion of the central element is constructed of a flexible material. 9. A multi resonant radio transceiver operable on either of two frequency bands, including an antenna arrangement for transmission and reception and comprising a tubular element and a central element; wherein the tubular element is electrically connected to a feed element by way of a conductive base portion and has a rim defining a distal portion of the tubular element; wherein the central element extends from the base portion, within the tubular element and outwardly from the rim of the tubular element; wherein, in use, at the higher frequency, the tubular element interacts with the central element to provide a short circuit whereby the antenna resonates between the rim of the tubular element and the feed; and at the lower frequency, the antenna operates in a fundamental mode of operation over the whole of the antenna length. 10. A multi resonant radio antenna operable at two wavelengths i and A 2, where A i corresponds to the higher frequency, comprising a first monopole element having a length corresponding to a quarter of the wavelength of the lower operating frequency and a cylindrical element comprising a coaxial stub which surrounds the first monopole element having an electrical length corresponding to λ 1/4 the cylindrical portion being electrically connected at the top of the monopole, providing the monopole with an exposed λ 1/4 length between the stub and a base of the antenna; wherein at the lower operating frequency, current is induced on the outer surface of the coaxial stub which is in phase with the exposed A 1/4 section and the antenna performs as a quarter wave monopole; and wherein at the higher operating frequency the stub presents a high impedance at the top of the monopole whereby the effective length of the monopole is 1/4, as measured from the base of the antenna. 1 1. A method of operating a multi resonant radio antenna operable at two wavelengths A and A 2, where A i corresponds to the higher frequency, the antenna comprising a first monopole element having a length corresponding to a quarter of the wavelength of the lower operating frequency and a cylindrical element comprising a coaxial stub which surrounds the first monopole element having an electrical length corresponding to λ 1/4 the cylindrical portion being electrically connected at the top of the monopole, providing the monopole with an exposed λ ι/4 length between the stub and a base of the antenna; wherein at the lower operating frequency, current is induced on the outer surface of the coaxial stub which is in phase with the exposed ι/4 section and the antenna performs as a quarter wave monopole; and wherein at the higher operating frequency the stub presents a high impedance at the top of the monopole whereby the effective length of the monopole is /4, as measured from the base of the antenna. 12. A multi resonant radio antenna according to claim 1 , comprising first and second conductive elements which antenna elements extend relative to a ground plane; wherein the elements of the antenna structure are adapted to couple between themselves to provide a variable phase velocity for surface currents of the radio signals. 13. A multi resonant radio antenna according to claim 1 , comprising first and second coupled lines operable to transmit and receive radio signals, a feed and a ground plane; wherein the feed exchanges radio signals with the first line, which line extends relative to the ground plane, and; wherein the lines are coupled such that at a first frequency, the phase velocity of the surface currents across the coupled lines are increased and, at a second, higher frequency, the phase velocity of the surface currents across the coupled lines are decreased. 14. A multi resonant radio antenna according to claim 13 wherein the coupled transmission lines are coupled at a distal end of . the first conductive element. 15. A multi resonant radio antenna according to claim 13, wherein said second line is parallel with said first line. 16. A multi resonant radio antenna according to claim 13, further comprising a third coupled line. 17. A multi resonant radio antenna according to claim 13, further comprising a third coupled line, which third line is parallel with said first line. 18. A multi resonant radio antenna according to claim 1 , comprising first, second and third coupled lines operable to transmit and receive radio signals, a feed and a ground plane; wherein the feed exchanges radio signals with the first line, which line extends relative to the ground plane, and; wherein the lines are coupled such that at a first frequency, the phase velocity of the surface currents across the coupled lines are increased and, at a second, higher frequency, the phase velocity of the surface currents across the coupled lines are decreased. 19. A multi resonant radio antenna according to claim 1 , comprising adjacently placed conductive lines, which lines have a frequency response whereby, at a high frequency mode of operation, the phase velocity of surface currents is reduced and at a lower frequency mode of operation, the phase velocity of surface currents is increased. 20. A method of operating a multi resonant radio antenna, said multi-resonant antenna comprising first and second coupled lines operable to transmit radio signals, a feed and a ground plane, wherein the first line extends relative to the ground plane; wherein, in a transmit mode, the method comprises the steps of providing radio signals, via the feed, to the first line, wherein the lines are coupled such that at a first frequency, the phase velocity of the surface currents across the coupled lines are increased and, at a second, higher frequency, the phase velocity of the surface currents across the coupled lines are decreased; whereby the lines resonate and the signals are transmitted via the lines. 21. A method of operating a multi resonant radio antenna, said multi-resonant antenna comprising first and second coupled lines operable to receive radio signals, a feed and a ground plane, wherein the first line extends relative to the ground plane; wherein, in a receive mode, the method comprises the steps of receiving radio signals, via the coupled lines such that at a first frequency, the phase velocity of the surface currents across the coupled lines are increased and, at a second, higher frequency, the phase velocity of the surface currents across the coupled lines are decreased; whereby the lines resonate and the coupled radio signals are output via the feed.

Inventors: JOHNSTON RONALD HARVEY, KITCHENER DEAN, ROBSON JULIUS GEORGE, JOHNSTON, RONALD, HARVEY, KITCHENER, DEAN, ROBSON, JULIUS, GEORGE
Applicants: NORTHERN TELECOM LTD, NORTHERN TELECOM LIMITED
Application year: 1997
Main classification: H01Q5
IPC codes: H01Q1, H01Q5, H01Q9, H01Q21