Vibration detection

Abstract: It is often desirable to detect the vibrations of an object without making physical contact with it, and of the several devices and procedures which will allow this many involve reflecting shortwave electromagnetic energy, typically microwaves and laser light, off the vibrating surface. For conventional such devices to work the displacement of the vibrating surface must be significant compared to a wavelength of the radiation used to detect them. A good example of a vibrating object is a water distribution pipe buried below ground; such pipes vibrate (to produce a typical "hissing" sound) when they leak, and it would be desirable to be able to to detect leaks by sensing this vibration in the manner described above. Unfortunately, the ground absorbs short-wavelength radiation very rapidly, and neither laser beams nor microwave radiation penetrate to the depth required. Longer wavelength radar signals can penetrate this far, but using such long wavelength radiation makes vibrations harder to detect because the physical displacements produced are far smaller than the wavelength, and the effect of vibration on the reflection is unmeasurable by conventional means. The present invention suggests how a radar can be used. It proposes employing a radar system in which a signal (2,28) equivalent to the echo from the object (6) when it is not vibrating is subtracted (33) from the actual returning echo signal (7,9) before that echo signal is detected by the system's receiver (10), so that the signal then actually received by the receiver consists only of that very much smaller component of the echo which is due to the displacements of the vibrating object from its rest position. This component may then be amplified, and the displacements may then be estimated from it.

Description: This invention is concerned with vibration detection, and relates in particular to the detection of the vibrations of an object by means of radar. There are many situations in which it is desirable to detect the vibrations of an object without making physical contact with it. This may be because the object is a lightweight structure, and physical contact might disturb its vibrations and prevent their being measured in the undisturbed state, or it may be because the object is difficult to make contact with - as is the case where the object is a pipe buried underground. There are many devices and procedures which will allow the non-contact detection and measurement of vibration. Many of these rely on short-wave electromagnetic radiation, including microwaves and laser beams, and a good example of one such is the laser velocimeter marketed by Ometron Ltd of the UK. This device shines a laser beam onto the surface of the object whose vibration is to be measured, detects the reflection of the beam, and is then able to deduce the velocity of vibration of the surface of the object at the point of reflection by means which involve interference between the reflected beam and a reference beam from which the transmitted beam was derived. For conventional devices to work, the displacement of the vibrating surface must be significant compared to a wavelength of the radiation used to detect them. If the vibrations are substantially smaller then these conventional means of detection will not be effective. However, there are some situations in which short-wavelength radiation cannot be reflected from the object, and a good example is that of water distribution pipes buried one or two metres (3-7 feet) below ground. In the neighbourhood of a leak in such a pipe the typical "hissing" sound of the leak is conducted along the pipe as vibration. This can be heard quite some distance away, but it would be desirable to be able to use more up-to-date equipment to detect water leaks by sensing this vibration in the manner described above. Unfortunately, the ground absorbs short-wavelength radiation very rapidly indeed, and neither laser beams (with wavelengths of the order of micrometres) nor microwave radiation (with wave lengths of the order of millimetres) would penetrate to the depth required. In a case such as this, then, radar signals - signals with much longer wavelengths (of the order of centimetres; often around 10 cm) - are preferable because they can penetrate up to several metres of ground, and can therefore be used to obtain a reflection from such buried pipes. Unfortunately, using radiations of these longer wavelengths makes vibrations much harder to detect, simply because the displacements produced by vibrations of things like pipes at audio frequencies are typically of the order of a micron (1 micrometre, or 10 -6 m) or less, i.e. they are far smaller than the radar wavelength, by a factor of a hundred thousand or so. The effect of vibration on the returning radar reflection (i.e. the echo from the vibrating target) is so small as to be to all intents and purposes quite unmeasurable by conventional means. Accordingly, this difficulty has hitherto prevented the use of radar for vibration detection. The present invention, however, suggests a way in which radar, with its relatively long wavelengths and thus sufficient ground penetration, can be used - a way in which the vibrational information carried by the echo from a vibrating object such as a pipe can be extracted to provide a useful output. More specifically, the invention proposes employing a radar system in which a signal equivalent to the echo from the object when it is not vibrating is subtracted from the actual returning echo signal before that echo signal is detected by the system's receiver, so that the signal then actually received by the receiver consists only of that very much smaller component of the echo which is due to the displacements of the vibrating object from its rest position. This component may then be amplified, and the said displacements may then be estimated from it. In one aspect, therefore, the invention provides a method of measuring small, time-varying displacements of a vibrating object, which method involves the object being illuminated with electromagnetic radiation of a radar nature - that is, of a wavelength considerably larger than the displacements - and the returned echo signal then being amplified to enable its subsequent use to provide useful data, in which method a signal equivalent to the echo from the object when it is not vibrating is subtracted from the returned echo signal before it is amplified so that the signal that is amplified consists only or mostly of the very much smaller component of the echo which is due to the displacements of the vibrating object from its rest position. In another aspect, the invention provides apparatus for measuring small, time-varying displacements of a vibrating object, which apparatus includes transmitting means for illuminating the object with electromagnetic radiation of a radar nature - that is, of a wavelength considerably larger than the displacements - and receiver means for receiving and amplifying the returned echo signal to enable its subsequent use to provide useful data, which apparatus also includes: generating means to supply a second signal equivalent to the echo from the object when it is not vibrating; and coupler means to subtract that second signal from the returned echo signal before it is amplified so as to provide a third signal that consists only or mostly of the very much smaller component of the echo signal which is due to the displacements of the vibrating object from its rest position, and then to feed that third signal to the receiver amplifier. The invention provides a method of, and apparatus for, measuring small, time-varying displacements of a vibrating object. Its primary purpose is for measuring vibrations in underground objects, specifically pipes (such as water pipes) that may leak their contents, and that will generate vibrations, manifested as a hissing noise, when they do so. Most typical of this is a mains water pipe. The invention involves the object being illuminated, by a transmitter thereof, with electromagnetic radiation of a radar nature - that is, of a wavelength considerably larger than the displacements. In this context, "considerably larger" means in general four or even more orders of magnitude larger - as noted above, a leaking mains water pipe, for example, produces audio frequency vibrations with amplitudes typically of the order of a micron (1 micrometre, or one hundredth of a centimetre [10 -6 m]) or less; a typical conventional "centimetre" radar has a wavelength a hundred thousand times that. Illuminating the object with the radar signal requires a generator producing the correct signal and feeding it to a transmitting antenna. Moreover, to receive the echo signal there is required a receiving antenna suitably coupled to the receiver proper (which will include appropriate amplifying and other stages, as needed). This is in principle all quite well known, and needs no further comment here. Nevertheless, the following comments may be helpful for a fuller appreciation of the invention. Firstly, the signal from the receiver antenna is preferably demodulated by mixing with some of the transmitter output signal. Secondly, the transmitted signal is conveniently continuous, rather than being in the form of pulses of finite duration. Thirdly, most preferably the transmitter and receiver antennae are separate and distinct, and direct coupling between them is caused to be small or preferably negligible by suitable choice of their relative positions. This latter may be more easily achieved if the transmitter and receiver antennae are dipole antennae, and are disposed at right angles to each other. In the invention the returned echo signal is amplified to enable its subsequent use to provide useful data, but before it is amplified a signal equivalent to the echo from the object when it is not vibrating is subtracted therefrom, leaving only (or mostly) the vibrational component by itself. This subtraction from the returned echo signal takes place before the returned echo signal is amplified; most conveniently, then, it is effected immediately after the returned signal has been picked up by the receiving antenna, and so before the antenna's output is fed into the receiver proper. The signal to be subtracted from the returned echo signal is a signal equivalent to the echo from the object when it is not vibrating; such a signal is, of course, a very small (low amplitude) version of the radiated signal used to illuminate the object in the first place, which radiated signal is generated in the equipment's transmitter. Thus, to provide the required signal equivalent to the echo from the object when it is not vibrating there need merely be taken a suitably-small part of the generated signal, which can then be subtracted from (by suitably modifying its phase and then electronically adding it to) the returned echo signal. This small part can notionally be provided by feeding a portion of the generated signal to an attenuator. However, plain attenuation may not be sufficient, and therefore it is preferred to form the small signal using circuitry that enables the signal to be modified in amplitude and phase in exactly the manner required. Such circuitry is a network device of the type discussed in more detail hereinafter with reference to the accompanying Drawings; it allows both the gain and phase of its output signal to be changed relative to its input signal in accordance with operator-adjusted controls. In the hereinafter-described embodiment of such a network device, the input signal is passed to a variable-gain amplifier and a quadrature filter, and the output from the filter is fed to another variable-gain amplifier (either amplifier may have a gain which is negative, zero, or positive depending on its adjustment). The amplifiers' output signals are then passed to a summing filter the output of which is the sum of the two amplifier outputs and constitutes the output of the network device. By adjusting the gains of the amplifiers the network device output signal may be caused to have any desired amplitude and phase relative to the original input signal. The network device can be automatically adjusted using an active control system. A convenient form of such a device is in essence a feedback circuit which acts to minimise the time-average of the squared amplitude of the returned signal, and preferably operates according to the method of gradient descent (so that the circuit detects the amplitude of the echo received by the antenna and repeatedly modifies the amplitude and phase of the signal subtracted therefrom in such a way as to reduce the time-average of the signal's squared amplitude as far as possible). In the invention the signal equivalent to the echo from the object when it is not vibrating is subtracted from the returned echo signal before it is amplified so that the signal that is amplified consists only or mostly of the very much smaller component of the echo which is due to the displacements of the vibrating object from its rest position. This "subtraction" is most conveniently effected by feeding two signals to the two inputs of a directional coupler (using directional couplers to combine one RF signal with another is a well-known technique in RF systems engineering, and needs no detailed explanation here; in the Specific Description hereinafter with reference to the accompanying Drawings there is described one arrangement wherein the relative phases of the two signals are adjusted so that they may be added by a direction coupler to give the required result). It will be seen that the invention provides a means of measuring small, time-varying displacements of an object using a radar signal whose wavelength is many orders of magnitude larger than the said displacements. A radar system is used in which a signal equivalent to the echo from the said object when it is not vibrating is subtracted from the returning signal before it is detected by the receiver of the radar system so that the signal received by the radar receiver consists only of the very much smaller component of the echo which is due to the displacements of the vibrating object from its rest position. This component may then be amplified and the said displacements may be estimated from it. Two embodiments of the invention are now described, though by way of illustration only, with reference to the accompanying diagrammatic Drawings in which: Figure 1 shows a block diagram for a vibration-detecting and -measuring radar system of the invention; Figure 2 shows a block diagram of a network device for producing a subtraction signal; and Figure 3 shows the block diagram for an alternative configuration of the radar system. Figure 1 is a block diagram of a radar system of the invention. It shows the signal paths between various components of the radar system. For the most part the components are standard RF (radio frequency) components, and they are to be connected together using conventional techniques which will be well-known to anyone practised in the art of constructing RF systems which use frequencies in the order of 500 megahertz. The circuit is now explained, as follows. A crystal oscillator (1) produces a signal (2) at a fixed and stable angular frequency ω 0 , and with constant amplitude A 2 . The signal 2 is amplified by an amplifier (3) which feeds a continuous signal (30) to a transmitter antenna (4). The antenna 4 consequently radiates a beam of radio waves (5). If this beam 5 encounters a reflector (6, such as a buried pipe) then a reflected beam (7) may return to the receiver antenna (8). The antenna 8 should preferably be positioned so that direct coupling between it and the transmitter antenna 4 is small (and preferably negligible), so that in the absence of any reflecting object 6 the signal at the receiving antenna 8 will also be small (or negligible). For example, if the antennae 4,8 are simple dipole antennae then they may be placed at right angles to each other. The antenna 8 passes the received signal (9) to an amplifier (10) via a coupler (33) which is described further hereinafter. The amplified signal (11) is passed to two identical mixers (12,13; a mixer is a device with two input signals whose output signal is the arithmetic product of its two input signals). In the first mixer 12 the signal 11 is mixed with (i.e. multiplied by) the oscillator output signal 2, while in the second mixer 13 it is mixed with the output of a quadrature filter (15) which is itself driven by the oscillator output signal 2. This results in two signals (16,17) output by the mixers 12,13 respectively. The signals 16,17 will now be described in more detail by means of the following Equations. Let the signal 2 from the oscillator 1 be denoted by the symbol s 2 (t), and let it be described by the Equation s 2 ( t ) = A 2 cosω 0 t where t is time. Let the output (14) of the quadrature filter 15 be denoted by the symbol s 14 ( t ). Let its output be described by the Equation <fax> s 14 ( t ) = A 14 sinω 0 t i.e. the signal 14 is to be of the same amplitude as the signal 2 but is to be phase shifted by 90° with respect to signal 2. Let the signal 11 be denoted by the symbol s 11 ( t ), and let it be described by the Equation s 11 ( t ) = A 11 cos(ω 0 +  11 ) where  11 represents the phase shift of the reflected signal 11 with respect to the oscillator output signal 2. Let the signal (16) be denoted by the symbol s 16 ( t ), and let the signal (17) be denoted by the symbol s 17( t ). Then the output signals 16,17 of the mixers will be described by the Equations These Equations demonstrate that the output signals 16,17 from the mixers 12,13 each contain a component which depends on the phase  11 of the reflected signal 11, and is independent of the oscillator frequency ω 0 - i.e., it is unmodulated and a second component which is modulated at twice the oscillator frequency. These signals 16,17 are then passed through low-pass filters (18,19) which are adjusted so that they remove the component which is modulated at twice the oscillator frequency while passing the unmodulated component. The output signals [24; to be denoted s 24 ( t ), and 25 (to be denoted s 25 ( t )] are therefore given by the Equations s 24 ( t ) = ½ A 2 A 11 cos 11 and s 25 ( t ) = ½ A 2 A 11 sin 11 The phase of the received signal 9 may therefore be deduced from the signals 24,25 via the Equation  11 = arc tan ( ( s 25 ( t )) ( s 24 ( t )) ) The phase  11 of the received signal 9 is to be used as an indicator of small displacements of the body 6. If the body 6 moves, then the length of the radio wave propagation path from transmitter antenna 4 to receiver antenna 8 via the body 6 changes and consequently the phase of the received signal 11 also changes. Let the velocity of propagation of the radio waves 5,7 be c. If a change δ 11 in the phase  11 is detected then the change δ/ in path length to which this corresponds is given by δ 1 = c ω 0 δ 11 This Equation may also be used to obtain an order of magnitude estimate of the size of the phase shift which may result from a displacement of the object which causes a change δ/in the path length in the case of a ground-probing radar which is to be used to detect vibrations in buried pipes. Vibrational displacements may be expected to be of order 10 -6 m. The angular frequency ω 0 cannot be much higher than order 10 9 rad/sec because of absorption by the ground. If c is taken to be about 5x10 7 m/sec then changes in phase cannot be expected to exceed order 2x10 -5 radians. This is too small to be detected by conventional means, and the following means may be used instead. Equation (3) gives the received signal including the effects of the phase shift  11 . We now expand this to allow for the case where  1 varies to a small extent with time by an amount δ 11 ( t ) which is the phase shift due to vibrations of the object 6 and which is itself a function of time t. We write  11 = ( 11 ) 0 + δ 11 ( t ) where ( 11 ) 0 is the constant component of  11 . We may now write a more general form of Equation 3: s 11 ( t ) = A 11 cos(ω 0 t + ( 11 ) 0 + δ 11 ( t )) Given that the phase shift δ 11 ( t ) is expected to be very small we may expand Equation (14) and obtain the following very accurate approximation to the received signal 11: s 11 ( t ) = A 11 { cos ((ω 0 t +( 11 ) 0 )-δ 11 ( t ) sin (ω 0 t +( 11 ) 0 ))} This Equation expresses s 11 ( t ) as the sum of an time-periodic, constant-amplitude term (i.e., the first term in the curly brackets hereinafter to be referred to as the non-varying part of s 11 ( t )) and a much smaller time-periodic, varying-amplitude term (i.e., the second term in the curly brackets hereinafter to be referred to as the varying part of s 11 ( t )). The first term is equivalent to the signal that would be received from the object if it were not vibrating; the second is due entirely to its vibrations. The signal s 11 ( t ) will be dominated by the first term within the curly brackets because the term δ 11 ( t ) is so small in comparison to unity. This would normally render the second term undetectable because the gain of the receiver amplifier 10 will be set to accommodate the signal s 11 ( t ) as a whole in order to avoid saturating the receiver, and the second term will then be so small that it will be masked by noise generated within the radar circuitry itself. To avoid this we arrange to cancel out the part of the signal s 11 ( t ) represented by the first term before the signal reaches the amplifier 10 so that the gain can be set to accommodate the second term only. This cancellation may be effected as follows. Referring to Figure 1, a secondary signal (28) is added to the signal 9 passing to the receiver amplifier 10 by means of a directional coupler (33: a directional coupler is a standard reactive RF device which can be used to produce the sum of two input signals at its output). The signal 28 fed to the directional coupler 33 is derived from the oscillator 1. The amplitude and phase of this signal 28 are arranged to be such that the first term in the curly brackets in Equation 14 is cancelled so that the signal 9 reaching the amplifier 10 consists only of the second term in the curly brackets in Equation 14. This may be accomplished by the following procedure. First of all the secondary signal 28 is turned off and the transmitter signal 30 is turned on. Resulting values of signals 24,25 are measured. From these a value of  11 may be deduced by means of Equation 11.  11 is measured many times consecutively by this means, and the values are averaged so that the time-varying component is suppressed and a value is obtained which corresponds closely to the quantity ( 11 ) 0 in Equation 15. The amplitude A 11 is next deduced by the use of the following Equation derived from Equations 9 and 10, and from a knowledge of the amplitude A 2 of the oscillator signal: A 11 = 2 ( s 24 ( t ) 2 + s 25 ( t ) 2 ) 4 A 2 We denote these values for ( 11 ) 0 and A 11 by ( 11 ) T and ( A 11 ) T respectively. The transmitter signal 30 is then turned off, and a test signal is supplied to the directional coupler 33 via the network device (22). Network device 22 changes the gain and phase of its output signal relative to its input signal in accordance with two controls adjusted by the operator (its operation is explained in more detail hereinafter). The gain and phase shift of network device 22 are set to standard values (preferably unity and zero respectively), and the above procedure of deducing the values of  11 and A 11 is repeated. The new values for  11 and A 11 we denote by ( 11 ) C and ( A 11 ) C respectively. The controls of network device 22 are then adjusted until a condition is obtained where ( 11 ) C = ( 11 ) T + Π and ( A 11 ) C = ( A 11 ) T i.e. the directional coupler now produces a receiver signal 9 which is identical to the invariant component of the signal produced by the reflected beam 7 except that it has the opposite polarity. Since the receiver system is linear, this implies that the superposition of the two signals will lead to cancellation of the non-varying component in Equation 15. Finally, both signals 28 and 30 are turned on simultaneously. The application of the signal 28 will cancel the invariant part of the reflected beam 7 at the receiver, and the signal 11 will now be given by the Equation s 11 ( t ) = - A 11 {δ 11 ( t )sin(ω 0 t + ( 11 ) 0 )} By repeating the argument steps represented by Equations 4 to 8 it can be shown that the output signals 24,25 are now given by the Equations s 24 ( t ) = ½ A 2 A 11 δ 11 ( t )sin( 11 ) 0 ) and s 25 ( t ) = ½ A 2 A 11 δ 11 ( t )cos( 11 ) 0 ) so that s 24 ( t ) and s 25 ( t ) are seen to be proportional to δ 11 ( t )) and the desired indicator of displacements of the body 6 may be obtained by application of the following Equation to the signal s 24 : δ 11 ( t ) = 2 s 24 ( t ) A 2 A 11 sin( 11 ) 0 or, if the phase of ( 11 ) 0 is such that its sin is zero, then the following Equation may be used instead: δ 11 ( t ) = 2 s 25 ( t ) A 2 A 11 cos( 11 ) 0 The network device 22 will now be explained. Figure 2 shows a block diagram of the network device 22. The input signal 2 passes to a variable-gain amplifier (41) and a quadrature filter (42). The output signal (43) from the filter 42 passes to a variable-gain amplifier (44). The amplifiers 41,44 may have gains which are negative, zero, or positive depending on their adjustment. The output signals 45,46 from the amplifiers 41,44 pass to a summing filter (47). The output 28 from the summing filter 47 is the sum of the two signals 45,46, and constitutes the output of the network device 22. By adjusting the gains of the amplifiers 41,44 the output signal 28 may be caused to have any desired amplitude and phase relative to the input signal 2. A second embodiment of the invention is now described (with reference to Figure 3); in this the network device 22 is controlled by an active control system which automatically adjusts the gains in the network so as to fulfil the conditions represented by Equations 17,18, thereby to cancel the unwanted part of the received signal 11 (the construction and operation of such active control systems is in itself well known to those versed in the art of active control). Figure 3 shows a block diagram of the second embodiment of the invention; the embodiment is substantially identical to that shown in Figure 1, but with the addition of the active control system (51). This system has two inputs which are the signals 24,25, and two output signals 52,53 which control the gains of the network device 22. The active control system 51 acts to minimise the sum of squares of its two input signals averaged over a period of a few seconds by varying its two output signals until a minimum is found. This may be implemented by using, for example, the gradient descent technique. The active control system 51 thereby acts in exactly the manner prescribed in the first embodiment of the invention to cancel the non-varying component of the received signal 11. The system will adapt itself to any change in the environment which causes changes in the quantities ( A 11 ) T and ( 11 ) 0 , such as moving the radar system. It will thereby save much labour in attaining and maintaining the desired cancellation of the non-varying component of the received signal 11.

Claims: A method of measuring small, time-varying displacements of a vibrating object (6), which method involves the object (6) being illuminated with electromagnetic radiation (5) of a radar nature - that is, of a wavelength considerably larger than the displacements - and the returned echo signal (7,9) then being amplified to enable its subsequent use to provide useful data, which method is characterised in that a signal (28) equivalent to the echo from the object (6) when it is not vibrating is subtracted from the returned echo signal (7,9) before it is amplified so that the signal that is amplified consists only or mostly of the very much smaller component of the echo which is due to the displacements of the vibrating object (6) from its rest position. A method as claimed in Claim 1, in which the subtraction from the returned echo signal (7,9) takes place immediately after the returned signal (7) has been picked up by the receiving antenna (8), and so before the antenna's output (9) is fed into the receiver proper. A method as claimed in either of the preceding Claims, in which the signal (28) to be subtracted from the returned echo signal (9) is a suitably-small part of the generated signal (2) suitably modified in phase and then electronically added to the returned echo signal (7,9). A method as claimed in Claim 3, in which the small part of the generated signal (2) is provided by feeding a portion of the generated signal (2) to an attenuator (22), and in which the attenuator (22) enables the signal (2) to be modified in amplitude and phase. A method as claimed in Claim 4, in which in the attenuator (22) is a network device wherein the input signal (2) is passed to a variable-gain amplifier (41) and a quadrature filter (42), the output (43) from the filter (42) is fed to another variable-gain amplifier (44), and the amplifiers' output signals (45,46) are then passed to a summing filter (47) the output (28) of which is the sum of the two amplifier outputs (45,46) and constitutes the output of the network device (22). A method as claimed in Claim 5, in which automatic adjustment of the network device (22) is enabled by using an active control system in the form of a feedback circuit (51) which acts to minimise the time-average of the square of the amplitude of the returned signal, and in which the feedback circuit (51) operates according to the method of gradient descent, so that the circuit detects the amplitude of the received echo (7,9) and repeatedly modifies the amplitude and phase of the signal (28) subtracted therefrom in such a way as to reduce the time-average of the square of the signal's amplitude as far as possible. A method as claimed in any of the preceding Claims, in which the signal (28) equivalent to the echo from the object (6) when it is not vibrating is subtracted from the returned echo signal (7,9) before it is amplified by using a directional coupler (33), and by feeding the former signal (28) to one input of the directional coupler (33) and the latter signal (9) to the other input thereof. A method as claimed in any of the preceding Claims, in which a first antenna (4) is used to transmit the radar signal (5) and a separate, second, antenna (8) is used to receive the echoes (7), and in which the transmitter and receiver antennae (4,8) are dipole antennae, and are disposed at right angles to each other. A method as claimed in any of the preceding Claims, in which the signal (9,11) received from the receiver antenna is demodulated by mixing with a signal (2) which is derived from the signal used to drive the transmitter antenna (4). Apparatus for measuring small, time-varying displacements of a vibrating object (6), which apparatus includes transmitting means (1,3,4) for illuminating the object (6) with electromagnetic radiation (5) of a radar nature - that is, of a wavelength considerably larger than the displacements - and receiver means (8,33,10) for receiving and amplifying the returned echo signal (7,9) to enable its subsequent use to provide useful data, which apparatus is characterised in that it also includes: generating means (22) to supply a second signal (28) equivalent to the echo from the object (6) when it is not vibrating; and coupler means (33) to subtract that second signal (28) from the returned echo signal (9) before it is amplified so as to provide a third signal that consists only or mostly of the very much smaller component of the echo signal (9) which is due to the displacements of the vibrating object (6) from its rest position, and then to feed that third signal to the receiver amplifier (10). Apparatus as claimed in Claim 10 when performing the method as defined in any of Claims 1 to 9.

Inventors: HARPER MARK FRANCIS LUCIEN, HARPER, MARK FRANCIS LUCIEN
Applicants: MECON LTD, MECON LIMITED
Application year: 1997
Main classification: G01S13
IPC codes: G01V1, G01H9, G01S7, G01S13