BALLAST SYSTEM

Abstract: An AC-AC ballast system for a discharge lamp (e.g., a fluorescent lamp), which includes a PFC converter which incorporates an isolation transformer, and a DC-AC inverter provided on the secondary side of the isolation transformer. Because the AC line input is isolated from the lamp load by the transformer in the PFC converter, the switching frequency of the PFC converter can be advantageously significantly higher than the lamp current frequency (and the switching frequency of the DC-AC inverter), to thereby enable a significant reduction in the size and weight of the ballast system, without an increase in the emission of EMI radiation from the lamp. In this regard, the ballast system of the present invention can be thought of as a two-frequency ballast system having an isolated PFC converter. In an exemplary embodiment of the ballast system of the present invention, the PFC converter includes a DC-DC converter, and a dither power factor correction circuit provided on the primary side of the isolation transformer, and the DC-AC inverter is a standard half-bridge or full-bridge DC-AC inverter. The PFC converter includes a first switching circuit and a first control circuit is provided for controlling the operation of the first switching circuit. In particular, the first control circuit operates the first switching circuit at a first switching frequency. The first control circuit also modulates the switching frequency of the first switching circuit in order to effect power factor correction. The DC-AC inverter includes a second switching circuit and a second control circuit is provided for controlling the operation of the second switching circuit. In particular, the second control circuit operates the second switching circuit at a second switching frequency which is much lower than the first switching frequency, e.g., at least ten times lower than the first switching frequency. The second control circuit also modulates the switching frequency of the second switching circuit in order toregulate the sinusoidal lamp current. Several alternative embodiments of the ballast system of the present invention are disclosed herein.

Description: Ballast system. The present invention relates to a ballast system for a lamp, comprising a AC-DC converter for generating a DC voltage out of a low frequency AC supply voltage, and a DC-AC converter for generating an AC voltage out of said DC voltage. Such a ballast system is known from EP 0323676. Generally the DC-AC converter can be a commutator such as a full bridge circuit switched at a low frequency. Very often, however, the DC-AC converter is a half bridge circuit generating a lamp current with a frequency in the order of 10 kHz. Electronic ballast lamps (EBLs) are in widespread use. In general, an EBL is a discharge lamp, e.g. , a fluorescent lamp, which is coupled to an electronic ballast circuit (system) which converts an AC line voltage into a high frequency AC output voltage for operating the lamp, and which utilizes a lamp current feedback signal to regulate the sinusoidal lamp current. With reference now to Fig. 1 , there can be seen a block diagram of a conventional electronic ballast system 20 which receives its power from a utility AC line 22, e.g. , from a standard 60 Hz residential outlet. The ballast system 20 includes an EMI filter 24 which filters out high-frequency noise from the ballast circuit. The AC power from the utility line is rectified by a rectifier 26, which produces a pulsating DC output. The pulsating DC output from the rectifier 26 is smoothed out by a high-frequency power factor correction (PFC) boost converter 28, which produces a smooth DC output with highly attenuated (i.e. , low percent) ripple. The PFC boost converter 28 functions to hold constant at zero the phase angle between the current and voltage waveforms of the pulsating DC output from the rectifier 26, to thereby provide a near-unity power factor (pf). In general, to meet industry requirements, a gas discharge lamp ballast should draw power from the power line with a power factor of at least 90% and harmonic distortion of less than 20% . The smooth DC output from the PFC boost converter 28 is then converted by a high-frequency DC-AC inverter 30 into a high-frequency (e.g. , 25-50 kHz) AC voltage which is delivered to the lamp 32 for ignition thereof. Since the input power of the system is relatively low frequency and the output power is relatively high frequency, a bulk capacitor C. is provided in the PFC boost converter 28 for energy storage, to thereby balance the input and output power. Isolation between the AC utility line input and the lamp load is provided by the inverter 30. A control circuit A is utilized to coarsely regulate the DC output from the PFC boost converter 28, and a control circuit B is utilized to control the operating frequency of the high-frequency DC-AC inverter 30, to thereby regulate the output power applied to the lamp 32. With reference now to Fig. 2, a typical embodiment of the conventional ballast system is depicted in partial schematic, partial block diagram form. As can be seen therein, the rectifier 26 is a full-bridge rectifier comprised of diodes D1-D4. The PFC boost converter 28 includes an inductor Ll connected in series with a forward-biased diode D, and a metal-oxide-semiconductor field-effect transistor (MOSFET) switch Q connected across the circuit. The control circuit A receives a voltage signal v and a current signal i indicative of the voltage and current values, respectively, of the pulsating DC output from the rectifier 26, at first and second inputs thereof, and receives a feedback signal from the output of the PFC boost converter 28 at a third input thereof. The control circuit A functions to selectively vary the ON duty ratio and/or the switching frequency of the switch Q in order to keep the voltage and current waveforms of the pulsating DC output from the rectifier 26 in phase with one another, and thus provide appropriate power factor correction. With continuing reference to Fig. 2, the DC-AC inverter 30 is a high- frequency half-bridge DC-AC inverter which includes a transformer T which isolates the lamp 32 from the AC line voltage. The high-frequency AC power produced by the DC-AC inverter 30 is delivered to the lamp 32 as a sinusoidal current through the L-C resonant circuit comprised of the inductor Lr and the capacitor Cr. The control circuit B receives a lamp current feedback signal and, in response thereto, controls the switching frequency of the MOSFET switches Ql and Q2 of the DC-AC inverter 30, to thereby regulate the high- frequency AC current delivered to the lamp 32. Since a fluorescent lamp acts as an antenna at high frequencies, the lamp current frequency is limited to about 100 kHz in order to prevent emission of excessive EMI radiation from the lamp. Typically, gas discharge lamps are operated at a frequency of 50 kHz. The conventional ballast system described above has at least one major shortcoming. Namely, the switching frequency of the DC-AC inverter is limited by the above-stated constraint on the lamp current frequency. This limitation on the switching frequency of the DC-AC inverter requires that magnetic components (e.g, inductors and isolation transformer), and other reactive elements (e.g. , capacitors) be designed for < 50- 100 kHz frequency, thereby imposing an unduly high lower limit on the size and weight of such components, thus unduly limiting the achievable miniaturization of the ballast system. The invention aims to provide an electronic ballast system for a lamp comprising an AC-DC converter which overcomes the above-described major shortcoming of conventional ballast systems. A ballast system as described in the opening paragraph is therefore according to the invention characterized in that the AC-DC converter comprises a power factor correction circuit equipped with inductive means; diode means coupled to said inductive means; first switching means coupled to said diode means for controlling a current through said inductive means; capacitive means coupled to said first switching means, and a first control circuit coupled to said first swithing means for rendering said first switching means alternately conductive and non-conductive at a high frequency f, and a DC-DC converter comprising second switching means coupled to the capacitive means; a high frequency transformer comprising a primary winding and a secondary winding, said primary winding being coupled to said capacitive means; rectifier means coupled to said secondary winding, and a second control circuit coupled to said first swithing means for rendering said first switching means alternately conductive and non-conductive at said high frequency f. The high frequency transformer functions as an isolation transformer. Since this transformer is part of the DC-DC converter and the AC line input is isolated from the lamp load by the transformer, the switching frequency f of both switching means in the AC-DC converter can be advantageously significantly higher than the lamp current frequency (and the switching frequency of the DC-AC inverter), to thereby enable a signficant reduction in the size and weight of the inductive means, the isolation transformer, and other reactive elements of the ballast circuitry, relative to conventional ballast systems, without an increase in the emission of EMI radiation from the lamp. In this regard, the ballast system of the present invention can be thought of as a two-frequency ballast system having an isolated power factor correction circuit. Preferably said first switching means comprises a first pair of switches and said second switching means compπses a second pair of switches. In an advantageous embodiment of a ballast system according to the invention compπsing relatively few components, said second switching means are formed by said first switching means and said second control circuit is formed by said first control circuit. To achieve a substantial miniaturization of said AC-DC converter, said high frequency f is at least ten times as high as the AC voltage generated by the DC-AC converter and preferably higher than 500 kHz. Good results have been obtained with embodiments wherein the power factor correction circuit comprises a dither power factor correction circuit, a half bπdge power factor correction converter or a push pull power factor correction converter. Miniaturization not only of the AC-DC-converter but also of the DC-AC converter can be realized in case the DC-AC converter is a high frequency pulse width modulated DC-AC converter compπsing third switching means for generating a high frequency voltage out of said DC voltage; a third control circuit coupled to said third switching means for generating a control signal for rendering said third switching means alternately conductive and non-conductive, said third control circuit comprising means for modulating the pulse width of said third control signal with a frequency equal to the frequency of the AC voltage generated by the DC-AC converter, and means for demodulating said high frequency voltage The frequency of the high frequency voltage is chosen relatively high. The modulation of the third control signal causes a modulation with the frequency of the AC voltage. The means for demodulating generate the AC voltage out of said modulated high frequency voltage. Since it is possible to choose the frequency of the third control signal much higher than the modulation frequency (= the frequency of the AC voltage generated by the DC-AC converter) the DC-AC converter can be miniaturized to to a substantive extent. Preferably said third switching means comprises a third pair of switches and said high frequency pulse width modulated DC-AC converter comprises a bridge circuit. These and various other features and advantages of the present invention will be readily understood with reference to the following detailed description taken in conjunction with the accompanying drawings, in which Fig 1 is a block diagram of a conventional electronic ballast system; Fig 2 is a partial schematic, partial block diagram of a typical embodiment of the conventional ballast system depicted in Fig. 1 ; Fig 3 is a block diagram of an electronic ballast system which constitutes a presently preferred embodiment of the present invention, Fig. 4 is a partial schematic, partial block diagram of an exemplary implementation of the ballast system of the present invention depicted in Fig. 3; Figs 5A-5D are schematic diagrams which depict successive stages of the operation of the the dither power factor correction circuit of the ballast system of the present invention depicted in Fig 4; Fig. 6 is the equivalent circuit for the DC-DC converter of the ballast system of the present invention depicted in Fig 4, Fig 7 is a diagram of the operating waveforms for the DC-DC converter of the ballast system of the preferred embodiment of the present invention depicted in Fig 6, Fig 8 is a partial schematic, partial block diagram of a first alternative embodiment of the ballast system of the present invention, Fig. 9 is a partial schematic, partial block diagram of a second alternative embodiment of the ballast system of the present invention, Fig 10 is a block diagram of a third alternative embodiment of the ballast system of the present invention, Fig. 1 1 is a partial schematic, partial block diagram of a first embodiment of the PWM DC-AC inverter of the ballast system of the third alternative embodiment of the present invention depicted in Fig. 10; Fig. 12 is a partial schematic, partial block diagram of a second embodiment of the PWM DC-AC inverter of the ballast system of the third alternative embodiment of the present invention depicted in Fig. 10; Fig. 13 is a diagram depicting the operating waveforms of the first embodiment of the PWM DC-AC inverter depicted in Fig. 1 1 ; and, Fig. 14 is a diagram depicting the operating waveforms of the second embodiment of the PWM DC-AC inverter depicted in Fig. 1 1 . Fig. 15 is a schematic diagram of a BIBRED converter which can be used in the ballast system of the present invention; and, Fig. 16 is a schematic diagram of a dither converter which can be used in the ballast system of the present invention. With reference now to Fig. 3, there can be seen a block diagram of an electronic ballast system 50 which constitutes a presently preferred embodiment of the present invention. As with the conventional ballast system described hereinabove, the ballast system 50 of the present invention includes an EMI filter 52 and a rectifier 56, e.g. , a half- bridge or full-bridge rectifier. However, rather than the non-isolated boost converter of the conventional ballast system, the ballast system 50 of the present invention includes a high- frequency PFC converter 58 which incoφorates an isolation transformer (not shown in Fig. 3) which isolates the AC utility line and the lamp 60. An energy storage capacitor Ce is connected across the PFC converter 58 on the primary side of the isolation transformer. The output of the isolated PFC converter 58 is a DC voltage which is well-regulated by the control circuit A. This well-regulated DC voltage is then inverted into a high-frequency AC voltage by a high-frequency DC-AC inverter 62, e.g. , a standard half-bridge or full-bridge AC-DC inverter or a half-bridge or full-bridge PWM AC-DC inverter. The high-frequency AC voltage produced by the DC-AC inverter 62 is delivered to the lamp 60 as a sinusoidal (AC) current for ignition thereof. The control circuit B regulates the sinusoidal lamp current. The switching frequency of the PFC converter 58 is significantly higher than the switching frequency of the DC-AC inverter 62, which is the same as the lamp current frequency. In this connection, the switching frequency of the PFC converter 58 is preferably > 500 kHz, whereas the switching frequency of the DC-AC inverter 62 and the lamp current frequency is preferably in the range of 25-50 kHz, in order to prevent excessive EMI radiation from the lamp 60, although the specific switching frequencies employed are not limiting to the present invention. With reference now to Fig. 4, there can be seen a partial schematic, partial block diagram of an exemplary implementation of the above-described presently preferred embodiment of the present invention. With this particular implementation, the PFC converter 58 is comprised of a dither power factor correction circuit 70 and a high-frequency DC-DC converter 72. The dither power factor correction circuit 70 includes an inductor Ll, diodes Dl and D2, MOSFET switches Ql and Q2, and energy storage capacitor Ce. The DC-DC converter 72 includes the MOSFET switches Ql and Q2, capacitors Cl and C2, transformer T, a full-bπdge rectifier comprised of the diodes D3-D6, and capacitors C3 and C4. The DC-AC inverter 62 is a half-bridge inverter comprised of the capacitors C3 and C4, MOSFET switches Q3 and Q4, and an L-C tank circuit comprised of an inductor Lr and a capacitor Cr connected across the lamp 60. With reference now to Figs. 5A-5D, the operation of the dither power factor correction circuit 70 will now be described With particular reference first to Fig. 5A, when the line voltage Vi is positive and the switch Ql is turned on, the hne voltage Vi is applied to the inductor Ll through the diode Dl and the switch Ql . The current through Ll rises linearly from zero to a positive peak value, and the energy is stored by the inductor Ll . With particular reference now to Fig 5B, when the line voltage Vi is positive and the switch Ql is turned off, the positive inductor current flows to the energy storage capacitor Ce through the body diode 76 of the switch Q2, to thereby transfer the energy stored in the inductor Ll when the switch Ql is turned on, to the energy storage capacitor Ce. The current through the inductor Ll falls linearly from its positive peak value to zero. With particular reference now to Fig. 5C, when the line voltage Vi is negative and the switch Q2 is turned on, the line voltage Vi is applied to the inductor Ll through the diode D2 and the switch Q2. The current through Ll rises linearly from zero to a negative peak value, and the energy is stored by the inductor Ll With particular reference to Fig. 5D, when the line voltage Vi is negative and the switch Q2 is turned off, the negative inductor current flows to the energy storage capacitor Ce through the body diode 78 of the switch Ql , to thereby transfer the energy stored in the inductor Ll when the switch Q2 is turned on, to the energy storage capacitor Ce. The current through the inductor Ll falls linearly from its negative peak value to zero. With reference again to Fig. 4, the control circuit A receives a full-wave rectified feedback signal from the output of the DC-DC converter 72, and modulates the switching frequency of the switches Ql and Q2 in such a manner as to effect power factor correction. The switches Ql and Q2 are turned on and off in complementary fashion, with a nominal 50% duty ratio, at high frequency, e.g. , > 500 kHz, so that the current through the inductor Ll is in discontinuous conduction mode (DCM). After the high-frequency components of the current through the inductor Ll are filtered by the EMI filter 52, the line current is a semi-sinusoidal wave in phase with the line voltage. The power factor is close to unity and the total harmonic distortion is low. Thus, the input power is controlled by the frequency modulation of the MOSFET switches Ql and Q2. With reference now to Fig. 6, there can be seen the equivalent circuit for the DC-DC converter 72, in which the secondary side is reflected to the primary side, and the output is replaced by a voltage source Vo. As can be seen, the transformer T has leakage inductances Llkl and Llk2, and a magnetizing inductance Lm. Before the switch Ql is turned on, the current ilk2 is negative. After the switch Ql is turned on, a positive voltage Vi is applied to the primary side of the transformer T, but a negative voltage Vo is applied to the secondary side of the transformer T through the diodes D4 and D6. The current ilkl flowing through Llkl and the current im flowing through Lm rise up. When ilkl exceeds im, the current ilk2 ( = ilkl-im) through Llk2 becomes positive and flows through diodes D3 and D5. When the switch Q2 is turned on, the operation is similar, but in reverse. The operating waveforms are depicted in Fig. 7. The DC-DC converter 72 can be designed to be operated with zero- voltage-switching in order to reduce switching losses and switching noise. In this connection, the duty ratio control signals A l , A2 applied by the control circuit A to the gates of the MOSFET switches Ql and Q2, respectively, are held inactive during a dead time, so that both switches Ql and Q2 are turned off duπng this dead time. During the dead time, the current ilkl through the leakage inductance Llk l is used to charge and discharge the drain- source capacitances Cdsl and Cds2 of the switches Ql and Q2, respectively. If the energy stored in the leakage inductance Llk l is not enough to effect complete charge and discharge of the drain-source capacitances Cdsl and Cds2, the energy stored in the magnetizing inductance continues to charge and discharge these capacitances. The DC-DC converter 72 can be designed for zero-voltage-switching over a full load and input voltage range. With reference again to Fig. 4, the high-frequency half-bridge DC-AC inverter 62 is conveniently the same as that in the conventional ballast circuit, except that the isolation transformer is eliminated, due to its inclusion in the PFC converter 58. The control circuit B receives a lamp current feedback signal and, in response thereto, modulates the switching frequency of the switches Q3 and Q4, in such a manner as to regulate the sinusoidal lamp current to a nearly constant rms value. In the preferred embodiment, the switches Q3 and Q4 are turned on and off in complementary fashion, with a nominal 50% duty ratio, at a high frequency, e.g. , 25-50 kHz. Thus, the output power is regulated by the frequency modulation of the switches Q3 and Q4. If the resonant frequency of the L-C circuit (Lr, Cr) is lower than the switching frequency of the DC-AC inverter 62, the inverter 62 can also be designed to be operated with zero- voltage-switching in order to reduce switching losses and switching noise. In this connection, the duty ratio control signals Bl , B2 applied by the control circuit B to the gates of the MOSFET switches Q3 and Q4, respectively, are held inactive during a specified dead time, so that both switches Q3 and Q4 are turned off during this dead time. With reference now to Fig. 8, there can be seen a first alternative embodiment of the ballast system of the present invention, which is the same as the exemplary implementation of the presently preferred embodiment depicted in Fig. 4, except that a high-frequency half-bridge PFC converter 80 is substituted for the high-frequency dither PFC converter 58. The half-bridge PFC converter 80 is comprised of a half-bridge power factor correction circuit 82 and a DC-DC converter 84. The half-bridge power factor correction circuit 82 includes a full-bridge rectifier comprised of diodes D 1-D4, an inductor Ll and diode D5, MOSFET switches Ql and Q2, and energy storage capacitor Ce. The DC- DC converter 84 is comprised of the MOSFET switches Ql and Q2, capacitors C l and C2, transformer T, a full-bridge rectifier comprised of diodes D6-D9, and capacitors C3 and C4. With reference now to Fig. 9, there can be seen a second alternative embodiment of the ballast system of the present invention, which is the same as the exemplary implementation of the presently preferred embodiment depicted in Fig. 4, except that a high-frequency push-pull PFC converter 90 is substituted for the high-frequency dither PFC converter 58. The push-pull PFC converter 90 is comprised of a push-pull power factor correction circuit 92 and a DC-DC converter 94. The push-pull power factor correction circuit 92 includes an inductor L l and diode D5 connected in series between a node NI and a first terminal 96 of the primary winding of the transformer T, and an inductor L2 and diode D6 connected in series between the node N I and a second terminal 98 of the primary winding of the transformer T, in parallel with the inductor Ll and diode D5. The node NI is coupled to the output junction node N2 of a full-bridge rectifier comprised of diodes D1-D4. A first MOSFET switch Ql is connected between the first terminal 96 of the primary winding of the transformer T and bottom rail 99, and a second MOSFET switch Q2 is connected between the second terminal 98 of the primary winding of the transformer T and the bottom rail 99, in parallel with the first switch Ql . The energy storage capacitor Ce is connected between the center tap 100 of the transformer T and the bottom rail 99. The remaining elements of the ballast system of this embodiment are the same as the corresponding elements of the ballast system of the presently preferred embodiment described hereinabove, and thus, a description thereof will be omitted. With reference now to Fig. 10, there can be seen a third alternative embodiment of the ballast system of the present invention, which is the same as the exemplary implementation of the presently preferred embodiment of the present invention depicted in Fig. 4, except that a high-frequency pulse-width modulated (PWM) DC-AC inverter 110 is substituted for the standard half-bridge DC-AC inverter 62. The PWM DC- AC inverter 1 10 may be a half-bridge PWM inverter as shown in Fig. 1 1 , or a full-bridge PWM inverter as shown in Fig. 12. The half-bridge PWM inverter works in the following manner. The pulse width of the duty ratio control signals Bl , B2 issued by the control circuit B are modulated by a relatively low-frequency signal (e.g. , a 25-50 kHz signal), resulting in a pulse-width modulated voltage Vab across points a and b. This pulse-width modulated voltage Vab is filtered by the L-C circuit comprised of the inductor Lo and the capacitor Co, to thereby produce a regulated, relatively low frequency output voltage (e.g. , 25-50 kHz AC voltage) for driving the lamp 60. The corner frequency of the L-C filter can be designed for operation at a frequency at least one decade higher than the modulation frequency, e.g. , 250 kHz, thereby enabling the size of the L-C filter to be significantly reduced. The switching frequency of the PWM inverter 110 (i.e. , the nominal frequency of the duty ratio control signals Bl , B2) can be at least one decade higher than the corner frequency of the L-C filter, e.g. , 2.5 MHz. During ignition of the lamp 60, the modulation frequency is on the same order as the corner frequency of the L-C filter so that a high-voltage output can be obtained to ignite the lamp 60, due to resonance of the L-C circuit. During steady-state operation of the lamp 60, the modulation frequency is kept to 25 kHz in order to prevent excessive EMI radiation from the lamp 60. Thus, the control circuit B regulates the lamp current by current mode control. The operating waveforms for the half-bridge PWM DC-AC inverter are depicted in Fig. 13, and the operating waveforms for the full-bridge PWM DC-AC inverter are depicted in Fig. 14. With reference now to Fig. 15, there can be seen a boost integrated with buck rectifier/energy storage/DC-DC converter (BIBRED converter) which constitutes an exemplary embodiment of a combined PFC/DC-DC converter (28, 42). The BIBRED converter includes an inductor 50 connected in series with a diode 52 and an energy storage capacitor 54, between the rectifier 26 and a first pole terminal 56 of the primary winding 58 of an isolation transformer 60, and a high-frequency switch 62 (e.g. , a MOSFET switch) connected between a node 64 intermediate the diode 52 and the capacitor 54, and a second pole terminal 66 of the primary winding 58 of the transformer 60. The anode of a diode 70 is connected to a first pole terminal 72 of the secondary winding 74 of the transformer 60, and a capacitor 76 is connected between the cathode of the diode 70 and a second pole terminal 78 of the secondary winding 74. An L-C filter comprised of an inductor 77 and a capacitor 75 is connected between a node 65 intermediate the capacitor 75 and the diode 70, and the first pole terminal 72 of the secondary winding 74 of the transformer 60. The switching frequency and duty ratio of the switch 62 are controlled by the control circuit B to generate a well-regulated DC current at the output of the BIBRED converter (28/42), at a very high switching frequency, e.g. , > 1 MHz. With reference now to Fig. 16, there can be seen a dither converter which constitutes another exemplary embodiment of a combined PFC/DC-DC converter (28, 42). The dither converter includes an isolation transformer 80 having a primary winding 82 which is center-tapped, and a secondary winding 84 which is also center-tapped. An inductor 86 is connected between the rectifier 26 and a first pole terminal 88 of the primary winding 82, and an energy storage capacitor 90 is connected between the center tap 92 of the primary winding 82 and the anode of a forward-biased diode 94 whose cathode is connected to a second pole terminal 96 of the primary winding 82. A high-frequency switch 97 (e.g. , a MOSFET switch) is provided on the pπmary side of the transformer 80. A half-bridge rectifier comprised of diodes 98 and 100, and an L-C filter circuit comprised of an inductor 102 and a capacitor 104, are provided on the secondary side of the transformer 80, with the capacitor 104 being coupled between the inductor 102 and the center tap 106 of the secondary winding 84. The switching frequency and duty ratio of the switch 97 are controlled by the control circuit B to generate a well-regulated DC current, at a very high switching frequency, e.g , > 1 MHz As will be readily appreciated by those skilled in the art, the operation or switching frequency of the DC-DC converter 42 (or the combined PFC converter 28/DC-DC converter 42) can be increased by employing any convenient zero-voltage-switching or soft¬ switching technique, e.g. , using a zero-voltage-switching quasi-resonant switch(es) or a zero- current-switching quasi-resonant switch(es), and/or by using a high-precision snubber circuit (not shown), in order to thereby further reduce the required size and weight of the transformer and other reactive components. Further, a half-bridge or other convenient type of commutator can be utilized instead of a full-bridge commutator. Moreover, the rectifier 26 and the PFC converter 28 can be eliminated if the input power is from a DC source, e.g. , a battery, rather than from an AC source such as a utility line.

Claims: CLAIMS: 1. A ballast system for a lamp, comprising a AC-DC converter for generating a DC voltage out of a low frequency AC supply voltage; a DC-AC converter for generating an AC voltage out of said DC voltage; characterized in that the AC-DC converter comprises a power factor correction circuit equipped with inductive means; diode means coupled to said inductive means; first switching means coupled to said diode means for controlling a current through said inductor; capacitive means coupled to said first switching means, and a first control circuit coupled to said first switching means for rendering said first switching means alternately conductive and non-conductive at a high frequency f, and a DC-DC converter comprising second switching means coupled to the capacitive means; a high frequency transformer comprising a primary winding and a secondary winding, said primary winding being coupled to said capacitive means; rectifier means coupled to said secondary winding, and a second control circuit coupled to said first switching means for rendering said first switching means alternately conductive and non¬ conductive at said high frequency f. 2. A ballast system as claimed in claim 1 , wherein said first switching means comprises a first pair of switches and said second switching means comprises a second pair of switches. 3. A ballast system as claimed in claim 1 or claim 2, wherein said second switching means are formed by said first switching means and said second control circuit is formed by said first control circuit. 4. A ballast system as claimed in one or more of the previous claims, wherein said high frequency f is at least ten times as high as the AC voltage generated by the DC-AC converter. 5. A ballast system as claimed in one or more of the previous claims, wherein said high frequency f is higher than 500 kHz. 6. A ballast system as claimed in one or more of the previous claims, wherein the power factor correction circuit comprises a dither power factor correction circuit. 7. A ballast system as claimed in claims 1-5, wherein said power factor correction circuit compπses a half bridge power factor correction converter. 8. A ballast system as claimed in claims 1-5 , wherein said power factor correction circuit comprises a push pull power factor correction converter. 9. A ballast system as claimed in one or more of the previous claims, wherein the DC-AC converter is a high frequency pulse width modulated DC-AC converter compπsing third switching means for generating a high frequency voltage out of said DC voltage; a third control circuit coupled to said third switching means for generating a control signal for rendering said third switching means alternately conductive and non-conductive, said third control circuit compπsing means for modulating the pulse width of said third control signal with a frequency equal to the frequency of the high frequency AC voltage generated by the DC- AC converter, and means for demodulating said high frequency voltage. 10. A ballast system as claimed in claim 9, wherein said third switching means comprises a third pair of switches and said high frequency pulse width modulated DC- AC converter comprises a bridge circuit

Inventors: GU WEN-JIAN, LIU RUI, GU, WEN-JIAN, LIU, RUI
Applicants: PHILIPS ELECTRONICS NV, PHILIPS NORDEN AB, PHILIPS ELECTRONICS N.V.
Application year: 1996
Main classification: H05B41
IPC codes: H02M1, H02M5, H05B41