Semiconductor memory device

Abstract: A dynamic semiconductor memory device according to the present invention comprises: at least first and second memory cell arrays having a plurality of memory cells selectively arranged at respective intersections of a plurality of word lines and a plurality of bit lines; a first sense amplifier section connected at an end of the first cell array to a plurality of bit line pairs formed by part of the plurality of bit lines of the first cell array, the plurality of bit line pairs having a folded bit line configuration; a second sense amplifier section connected to sets of bit line pairs, each formed by one of the remaining bit lines of the first cell array and one of part of the plurality of bit lines of the second cell array, the plurality of bit line pairs having an open bit line configuration; and a correction circuit for correcting the level of ease for reading data "0" and that of reading data "1".

Description: This invention relates to a dynamic semiconductor memory device and, more particularly, it relates to a dynamic semiconductor memory device designed to provide an improved sense amplifier system. In recent years, the developments in the technologies of ultrafine machining and the improvements in the memory cell structure have made it possible to produce, among others, remarkably highly integrated dynamic semiconductor memory devices having a 1-transistor/1-capacitor memory cell structure (hereinafter referred to as "DRAMs") so that less rigorous requirements may be applied to the arrangement of bit lines and word lines and the design of transistors. Currently, an open bit line configuration (hereinafter referred to as "open BL configuration") is used for memory devices up to 16K bits, whereas a folded bit line configuration (hereinafter referred to as "folded BL configuration") is used for devices between 16K bits and 64M bits, which represents the capacitance level of memory devices of the current generation. FIGS. 1A and 1B respectively show schematic illustrations of conventional open BL and folded BL configurations comprising sense amplifiers SA, word lines WL, bit lines BL and memory cells MC. In FIGS. 1A and 1B, reference symbol C BB denotes the coupling capacitance of a bit line BL. The open BL configuration is advantageous in that it can reduce the area occupied by the memory cell section of a memory device because all the intersections of the bit lines and the word lines can be used to arrange a memory cell, although it is accompanied by a difficult problem of arranging every sense amplifier within the width of a bit line because each bit line is extended for two cell arrays. The open BL configuration has an additional problem of rigorous design restrictions imposed on the sense amplifier section because of the fact that each pair of bit lines need a sense amplifier even when the sense amplifier is alternatively arranged among the cell arrays as shown in FIG. 1A (relaxed open BL configuration). On the other hand, in a semiconductor memory device of the folded BL configuration that has been in the mainstream from the generation of 64K-bit DRAMs up to the present, because memory cells can be arranged at only a half of the intersections of the bit lines and the word lines and a bit line pair is formed in a single cell array, a sense amplifier may be used for every four bit lines if sense amplifiers are arranged alternately at the oppositely disposed ends of the cell arrays as shown in FIG. 1B (a double folded BL configuration). Thus, the folded BL configuration can greatly reduce the pitch of arrangement of sense amplifiers and the sense amplifier section whose design rule is strict can be easily arranged, thereby the folded BL configuration is widely used for DRAMs. However, the folded BL configuration has a disadvantage that memory cells can be arranged at only a half of the intersections of the word lines and the bit lines so that the memory cell section of the device inevitably takes a large area as a whole to increase the dimensions of the final product or the chip. Particularly, it is extremely difficult to down-size the memory cell section of a dynamic semiconductor memory device having a folded BL configuration because of the rigorous design requirements imposed thereon if the device is of a 64M-bit type or of an above 256M-bit type. In short, known DRAMs face the problem of sense amplifier arrangement because of the rigorous requirements for the design rule of sense amplifier in an open BL configuration, whereas they face the problem of a large memory cell area and hence a large chip size in a folded BL configuration, although the design rule of the sense amplifier can be extremely relaxed. US-5,396,450 describes a dynamic random access memory device which describes an arrangement similar to that shown in Figure 2 in that it has a sense amplifier section including first and second sense amplifier circuits. Adjacent bit-line pairs of the bit lines include a first bit-line pair and a second bit-line pair, one of which has a folded bit-line arrangement and the other of which has an open bit-line arrangement. EP-A-0,240,155 describes a semiconductor memory device which has dummy cell capacitors connected between bit-lines and dummy word lines which are set to a potential between the high and low voltage supplies to reduce the potential difference between pairs of bit-lines. The inventors of the present invention have already proposed a dynamic semiconductor memory device having a reduced area occupied by the memory cell section if compared with a device with a folded BL configuration and, at the same time, subject to less rigorous requirements for the design of sense amplifier if compared with a device with the open BL configuration. In other words, the proposed dynamic semiconductor memory device is realized by optimally combining the open BL configuration and the folded BL configuration. For example, in a preferred embodiment of dynamic semiconductor memory device of the previously proposed invention comprising a plurality of cell arrays, the memory cells being selectively arranged on the intersections of word lines and bit lines, part of the bit lines of a first cell array form a plurality of bit line arrays and connected to a first sense amplifier at an end of a first cell array to realize a folded BL configuration. The remaining bit lines of the first cell array form pairs with corresponding bit lines of a second cell array located adjacent to the first cell array with a sense amplifier disposed therebetween to realize an open BL configuration. FIGS. 2 and 3A through 3C illustrate the configuration of the above arrangement and its operation. In another previously proposed invention, the dynamic semiconductor memory device comprising a plurality of cell arrays, the memory cells being selectively arranged on the intersections of word lines and bit lines, part of the bit lines of a first cell array form a plurality of bit line arrays and connected to a first sense amplifier at an end of a first cell array to realize a folded BL configuration. The remaining bit lines of the first cell array form pairs with corresponding bit lines on the reference side of the bit line pairs of the folded BL configuration to form a folded BL configuration before the sense amplifier starts operating, whereas they forms pairs with corresponding bit lines of a second cell array located adjacent to the first cell array with a sense amplifier disposed therebetween to realize an open BL configuration in a restoring operation of rewriting data in the memory cells after the start of the operation of the sense amplifier. FIGS. 4 and 5A through 5C illustrate the configuration of the above arrangement and its operation. Referring FIG. 2, assume, for example, that word line W A0 is selected. Then, the related memory data are read out onto BL 0 and BL 1 so that the data on BL 0 is read out onto the sense amplifier SA 0 when P01 is set to level "H" and SA 0 senses the data on BL 2 in the folded BL configuration, using as reference bit line BL 2 onto which no cell data are read out. The data on the BL 1 is read out onto SA 1 because P01 is at level "H" and SA 1 senses the data on BL 1 in the open BL configuration, using adjacently located BL 4 as reference bit line. If word line W A1 is selected, then, the folded BL configuration becomes applicable to BL 0 and BL 3 for sensing, whereas BL 1 and BL 4 are sensed in accordance with the open BL configuration. However, if word line W A2 is selected, then, P2 is brought to level "H" to read data onto BL 1 and the sense amplifier/cell array connection is changed in a manner as illustrated in FIGS. 3A through 3C for sensing with both the folded BL configuration and the open BL configuration. Referring now to FIG. 4 and assuming again that word line W A0 is selected, then, the related cell data are read out onto BL 1 and BL 2 and the data on BL 1 and BL 2 are read out to SA 0 and SA 1 respectively if  DE ,  D02 ,  CO and  CE are at level "H". Then, BL 0 becomes a reference bit line common to SA 0 and SA 1 to bring all of  DE ,  D02 ,  CO and  CE to level "L" so that the data are latched by SA 0 and SA 1 , which are then sensed according to the folded BL configuration. The SA 0 side data are then restored to BL 1 and BLD0 according to the folded BL configuration as  DE and  D02 are brought back to level "H". Now, since the data on the SA 1 side cannot be used as long as the data on the SA 0 side are being restored, they are restored according to the open BL configuration by using bit line BL 3 of the cell array located adjacent to BL 2 . If W A1 and W A2 are selected, then the sense amplifier/cell array connection is changed in a manner as illustrated in FIGS. 5A through 5C because of the difference of cell arrangement. However, the arrangement of FIGS. 2 and 4 are accompanied by the following problems. Namely, both the open BL configuration and the folded BL configuration are used for reading data with the embodiment of FIG. 2, whereas both the open BL configuration and the folded BL configuration are used for rewriting data with the embodiment of FIG. 4. It is well known that the open BL configuration and the folded BL configuration produce array noises with different levels and have different combination patterns of cell data "0" or "1" that determines the worst array noise level, different sense amplifier locations and different numbers and configurations of circuits for switches. With the conventional folded BL configuration, the difference between the writing voltage required for reading "0" and the writing required for reading "1" is compensated by using a technique of using dummy word lines for coupling. However, with the circuit arrangements of FIGS. 2 and 4, not only the voltage required for reading "0" and the voltage required for reading "1" have to be compensated but also there arises, as described above, a difference between the writing voltage required for reading "0" and the writing voltage required for reading "1" on the open BL configuration side and also between the voltage required for reading "0" and the voltage required for reading "1" on the folded BLs side. Thus, the performance of the arrays is determined by the poorer performance of reading "0" and that of reading "1". The above described differences may be also occurred when some of the switches are used. Additionally, for example, such differences may be also occurred when P2 in FIG. 2 is arranged close to the sense amplifier side relative to the P01. Still additionally, they may be also occurred when the bit line pairs in FIGS. 5A through 5C are formed by adjacently located lines or by those having another bit line disposed in-between as a result of selecting WL 0 , WL 1 , or WL 2 . On the other hand, with the circuit configuration of FIG. 4, while the number of memory cells is small than that obtainable from an open BL configuration, the cell area can be greatly reduced if compared with that obtainable from a conventional folded BL configuration. Additionally, sense amplifiers can be arranged along a direction perpendicular to the bit lines at a pitch much reduced than the pitch at which they are arranged with the open BL configuration, where only a single sense amplifier is allowed to be arranged for every two bit lines, although it is greater than the pitch at which they are arranged with the folded BL configuration, where only a sense amplifier may well be arranged for every four bit lines. In short, a dynamic semiconductor memory device according to the above invention is subjected to design restrictions that are more rigorous than those applied to the folded BL configuration but by far less rigorous than those applied to the open BL configuration. FIGS. 6A through 6C illustrate drives signals that can be used for a dynamic semiconductor memory device according to the above identified previously disclosed invention. Since a folded BL configuration is applied to the device for reading data, it is free from noises coming by way of word lines and specific to the open BL configuration and noises coming by way of unselected WLs can be eliminated to reduce the overall noise level as in the case of the folded BL configuration. However, inter-bit lines (hereinafter referred to inter BL-BL) noises that rise with the increase in the inter-bit line capacitance have been highlighted with the increase in the degree of integration of DRAMs. Inter BL-BL noises will now be discussed by referring to FIG. 4. FIG. 7 schematically illustrates a folded BL configuration to be used for data reading operation of the arrangement of FIG. 4. Looking at sense amplifier SA 0 , if the inter BL-BL capacitance is C BB and the reading signal is V S , BL 1 is subject to a noise with a level of +C BB V S coming from BL 0 while BL 2 is subject to a noise of -2C BB V S coming from both BL 1 AND BL 9 to make the overall maximum noise it receives equal to a level of 3C BB V S . Contrary to this, the maximum inter BL-BL noise of a conventional device with a folded BL configuration or a relaxed open BL configuration is equal to a level of 2C BB V S as seen from FIGS. 1A and 1B. With the configuration of FIG. 4, the level of the noises coming through the word lines, the plate and the substrate of the device that are specific to the open BL configuration are reduced because a folded BL configuration is adopted for reading data but the level of the inter BL-BL noise is one and a half times greater than that of the inter BL-BL noise of a conventional device with an open BL configuration. What is worse, since the reference bit lines can be positionally shifted depending on the positions of word lines WL of the device and a set of lines is formed by every three bit lines, the technique of reducing noise by dividing the bit lines by 2 n (n=natural number) and twisting the bit lines BL as shown in FIG. 7 cannot be applied to it. In view of the above identified problems, the inventors of the present invention have also proposed a method of noise reduction of twisting two of the three bit lines of each set of lines located at a position of 1/(3n). While noises can be reduced by forming 3n twists in cell arrays with the proposed method, the twists take an additional area to consequently increase the size of the finally produced memory chip. Additionally, with an open/folded hybrid BL configuration realized by combining an open BL configuration and a folded BL configuration in a manner as described above, the bit lines of the cell arrays and the sense amplifiers are connected in different ways depending on the addresses of the word lines. More specifically, the connection changes periodically for every three word lines as illustrated in FIG. 8. Assume now, for example, that word line WL 0 is selected, then control signal  0,1 goes to "H" to close the gate controlled by  0,1 . On the other hand,  2 goes to "L" to open the gate controlled by  2 . As a result, the bit lines, the memory cells and the sense amplifiers are connected in a manner as illustrated in FIG. 9A so that the data in memory cell M 1 is read out onto folded A and the data in memory cell M 2 is read out onto open sense amplifier B. If, on the other hand, word line WL 1 is selected, then the gate controlled by  0,1 is closed and the gate controlled by  2 is opened as shown in FIG. 9B so that, as a result, the data in memory cells M 3 and M 4 are read out onto folded sense amplifier A and open sense amplifier B respectively. If word line WL 2 is selected, then the gate controlled by  2 is closed and the gate controlled by  0,1 is opened as shown in FIG. 9C so that, as a result, the data in memory cells M 5 and M 6 are read out onto folded sense amplifier A and open sense amplifier B respectively. In reality, the word lines of the three different types are arranged cyclically so that control signal  for the connecting gates of the bit lines and the sense amplifiers cyclically changes for every three addresses of the word lines. Thus, a 3-phase address system is needed for controlling control signal . On the other hand, addresses that are used in a dynamic semiconductor memory device are typically expressed in binary digits of "0" and "1". Therefore, a word line address selected for controlling  0,1 and  2 and expressed in binary digits has to be converted. into a 3-phase address that represents one of three different word lines. If word lines WL 0 and WL 1 are selected, the gate of  0,1 is closed (to connect a bit line and a sense amplifier) and the gate of  2 is opened (to disconnect them). In other words, there is required a circuit that determines the remainder produced after dividing the address of a word line by three. FIG. 10 schematically illustrates known circuits that can carry out the above described conversion. Circuit A converts a two-digit binary number into a one-digit ternary number, whereas circuit B produces a ternary number equal to the digit of the lowest order of a ternary number obtained by adding two ternary numbers. Address input A i and /A i (i=1 to 7) are complementary input addresses. For example, if A i = "0", /A i ="1". When a word line is selected by address A i , the circuit produces outputs Z 0 , Z 1 and Z 2 as remainders. For example, if A 0 =A 1 =...=A 7 ="0" (/A 0 =/A 1 =...=/A 7 ="1"), then the outputs will be Z 0 ="1", Z 1 ="0", Z 2 ="0" and so on. Table 1 below illustrates the correspondence of inputs and outputs. A 7 A 6 A 5 A 4 A 3 A 2 A 1 A 0 Z 0 Z 1 Z 2 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 0 0 0 1 0 1 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 1 0 1 0 0 1 0 0 0 0 0 1 1 0 1 0 0 0 0 0 0 0 1 1 1 0 1 0 0 0 0 0 1 0 0 0 0 0 1 0 0 0 0 1 0 0 1 1 0 0 0 0 0 0 1 0 1 0 0 1 0 0 0 0 0 1 0 1 1 0 0 1 0 0 0 0 1 1 0 0 1 0 0 0 0 0 0 1 1 0 1 0 1 0 0 0 0 0 1 1 1 0 0 0 1 0 0 0 0 1 1 1 1 1 0 0 ... ... 1 1 1 1 1 1 1 1 0 1 0 As seen from the above table, when the remainder obtained by dividing an address by three is 0, then Z 0 =1, whereas Z 1 and Z 2 are equal to 1 when the remainder is equal to 1 or 2 respectively. On the other hand, the internal addresses of a dynamic semiconductor memory device are normally held unchanged regardless of the state of the external addresses when the device under a stand-by condition. Thus, when internal address A i =/A i ="0", circuit A produces outputs B 00 ="0", B 01 ="1" and B 02 ="1" and the last outputs Z 0 , Z 1 and Z 2 of Z 0 =Z 1 =Z 2 ="1". It should be noted here that Z i "1" (i=0, 1 or 2) represents that the remainder after dividing the address by three is equal to i and Z 0 =Z 1 =Z 2 ="1" represents a multiple selection. Thus, there exists a hazard of multiple selection in the selection of gate pi and dummy word line DWLi by means of surplus address Z i during a delay time after binary addresses A i , /A i are unequivocally defined and before surplus address Z i is unequivocally defined as the device moves from a stand-by state into an operating state. FIG. 11 illustrates such a hazard. As the device moves from a stand-by state into an operating state, there occurs a delay time after an address is unequivocally defined and decoded by the surplus circuit to unequivocally define a remainder or between an input operation and a corresponding output operation of the surplus circuit. The control signal being applied to the device may be inactivated in order to prevent a multiple selection from taking place before a surplus address is defined but the arrangement for such a configuration of inactivation can make the circuit complicated and the entire operation may become so much more time consuming because such a configuration inevitably involves a safety margin for the operation time. Thus, if a memory cell array shows more than one differences in the minimum writing voltage required for reading "0" and "1" in a conventional DRAM, the operating speed of the memory cell array is dominated by the worst performance for reading "0" and the worst performance for reading "1". Additionally, if a set of lines is formed by every three bit lines, the technique of twisting the bit lines to reduce interference noises among the bit lines cannot be applied to it. Still additionally, if two of the three bit lines of each set is twisted at a position of 1/3n, such twists take an additional area to consequently increase the size of the finally produced memory chip, although interference noises among the bit lines may be reduced. If internal addresses are decoded by means of a surplus circuit in a dynamic semiconductor memory device, there exists a hazard of multiple selection for the gates connecting the sense amplifiers and the bit lines and for the dummy word lines as the device moves from a stand-by state into an operating state. Therefore, it is an object of the present invention to provide a dynamic semiconductor memory device that has an improved safety margin for the operating time if a memory cell array shows more than one differences in the minimum writing voltage required for reading "0" and "1". It is another object of the present invention to provide a DRAM having a reduced memory cell area if compared with a comparable device with a folded BL configuration and subjected to less rigorous requirements for the design of sense amplifiers if compared with a device with an open BL configuration so that the DRAM may meet the demand for a reduced memory cell area and at the same time the demand for less rigorous requirements on the design of the sense amplifiers and for a reduced noise level among the bit lines. It is still another object of the present invention to provide a DRAM equipped with a surplus circuit that is free from the hazard of multiple section for the gates connecting the sense amplifiers and the bit lines and for the dummy word lines. According to a first aspect of the invention, there is provided a dynamic semiconductor memory device comprising: at least first and second memory cell arrays having a plurality of memory cells arranged at respective intersections of a plurality of word lines and a plurality of bit lines; a first sense amplifier section connected at an end of the first cell array to a plurality of first bit line pairs formed by part of said plurality of bit lines of said first cell array, said plurality of first bit line pairs having a folded bit line configuration; and a second sense amplifier section connected to a plurality of second bit line pairs, each of which is formed by one bit line of the remaining bit lines of said first cell array and one bit line of part of the plurality of bit lines of said second cell array, said plurality of second bit line pairs having an open bit line configuration; characterised in that said memory device further comprises shifting means adapted to correct imbalance between the read voltages of bit line pairs in an array, said shifting means comprising a plurality of control signal lines and a plurality of capacitors, wherein each one of the plurality of capacitors is associated with a respective bit line of the plurality of bit lines, each one of the plurality of capacitors having a terminal connected to its associated bit line and another terminal connected to one of the plurality of control signal lines, said shifting means shifting the read voltages on bit lines on which binary data is read out or on reference bit lines. The present invention also provides a dynamic semiconductor memory device comprising: at least first and second memory cell arrays having a plurality of memory cells arranged at respective intersections of a plurality of word lines and a plurality of bit lines; and a sense amplifier section connected to an end of the first cell array to a plurality of bit line pairs formed by part of said plurality of bit lines of said first cell array, wherein said plurality of bit line pairs have a folded bit line configuration, and the remaining bit lines of said first cell array form bit line pairs respectively with the reference bit lines of said bit line pairs having a folded bit line configuration to show a folded bit line configuration until the sense amplifiers start operating and then form bit line pairs having open bit line configuration respectively with part of the plurality of bit lines of a second cell array located adjacent to said first cell array with a sense amplifier section interposed between first and second memory cells arrays during a subsequent restoring operation which rewrites data into the memory cells after the operation of the sense amplifiers, characterised in that said memory device further comprises shifting means adapted to correct imbalance between the read voltages of bit line pairs in an array, said shifting means comprising a plurality of control signal lines and a plurality of capacitors, wherein each one of the plurality of capacitors is associated with a respective bit line of the plurality of bit lines, each one of the plurality of capacitors having a terminal connected to its associated bit line and another terminal connected to one of the plurality of control signal lines, said shifting means shifting the read voltages on bit lines on which binary data is read out or on reference bit lines. Preferable manners according to the first aspect of the invention are as follows: (1) The correction circuit performs at least either a capacitor coupling operation or an electric charge distributing operation for either the bit line or the reference bit line of a memory cell selected for reading data therefrom. (2) The correction circuit have two or more than two levels of ease of reading data. (3) Correction amounts corrected by the correction circuit are different between an open bit line structure and a folded bit line structure. (4) Correction amounts corrected by the correction circuit are different in types of selected word lines. (5) Two-thirds of the intersections of the plurality of word lines and the plurality of bit lines carry respective memory cells. (6) The bit lines having the folded bit line configuration and those having the open bit line configuration change positions depending on the positions of the selected word lines. (7) Two-thirds of the bit lines have a folded bit line configuration and the remaining one-third have an open bit line configuration. (8) A bit line with an open BL configuration is placed between each bit line pair with a folded BL configuration. (9) A set of lines is formed by every three bit lines, two of which form a bit line pair with a folded bit line configuration and the remaining bit line form an open bit line pair with a bit line of an adjacent cell array. A dynamic semiconductor memory device according to a first aspect of the invention is provided with a correction circuit for correcting each of the voltage differences between the level for reading data "1" necessary for cell writing voltage and the level for reading data "0" necessary for cell writing voltage if there are more than one such differences. By changing a correction amount of voltage by using the correction circuit for the above different types, the difference of correction amount between "0" reading and "1" reading can be eliminated. Thus, the area occupied by the memory cells of a dynamic semiconductor memory device according to the first aspect of the invention can be reduced than a conventional device with a folded BL configuration because the device of the invention combines an open BL configuration for arranging memory cells on all the intersections of the bit lines and the word lines and a folded BL configuration for arranging memory cells on half of the intersections of the bit lines and the word lines. Additionally, sense amplifiers can be arranged along a direction transversal to the bit lines at a pitch by far improved if compared with the pitch of sense amplifier arrangement of the open BL configuration that allows at most a sense amplifier to be arranged at every two bit lines, although less advantageous if compared with the open BL configuration with which a sense amplifier may be arranged at most at every four bit lines. The so-called relax open BL configuration with which a sense amplifier is arranged for every two bit lines is accompanied by a drawback of involving a large number of cell arrays if compared with the proper open BL configuration with which a sense amplifier is arranged for each bit line to tighten the pitch of sense amplifier arrangement. However, a dynamic semiconductor memory device according to the invention allows a pitch of sense amplifier arrangement less tight than that of the relax open BL configuration. In, short, the present invention exploits the most remarkable advantages of the conventional open BL configurations and those of the conventional folded BL configuration, overcoming the most remarkable drawbacks of those formats. As described above, according to the invention, by combining the open BL configuration and the folded BL configuration, there is provided a DRAM having a reduced memory cell area if compared with a comparable device with a folded BL configuration and subjected to less rigorous requirements for the design of sense amplifiers if compared with a device with an open BL configuration so that the DRAM may meet the demand for a reduced memory cell area and at the same time the demand for less rigorous requirements on the design of the sense amplifiers and for a reduced noise level among the bit lines. Such a device has the following additional advantages. (1) By arranging a correction circuit for correcting the difference between the cell writing voltage necessary for reading "0" and the cell writing voltage for reading "1", two or more than two differences of voltage that may exists between reading "1" and reading "0" in a cell array can be corrected to improve the safety margin of reading data from the chip. (2) By arranging a switching circuit for changing over the connection between the bit lines in the memory cell and the sense amplifier to make the bit line selected for reading a memory cell to be disposed between a bit line pair regardless of the position of the selected word line without resorting to techniques such a twist bit lines leading to a greater chip area, the level of interference noise among bit line can be reduced. Additionally, by arranging a circuit for shifting the potential of the bit line up and down by coupling two or more than two level of easiness of reading "1" and "0", any unbalanced easiness of reading "1" and "0" that may exist among bit pairs can be corrected. (3) By providing a dynamic semiconductor memory device having an open/folded hybrid BL configuration with a surplus circuit for controlling the operation of selecting gates connecting the sense amplifiers and the bit lines and the dummy word lines on the basis of cyclicity of the addresses of three word lines or by providing a circuit for holding the outputs of the surplus circuit to respective predetermined fixed potential levels regardless of the state of the first input signals when the memory cell arrays are in the stand-by state, the potential hazard of the control circuit when the chip is activated can be effectively avoided and a stable operation can be ensured for the device. This invention can be more fully understood from the following detailed description when taken in conjunction with the accompanying drawings, in which: FIGS. 1A and 1B are schematic diagrams respectively showing a conventional open BL configuration and a conventional folded BL configuration; FIG. 2 is a schematic diagram of a conventional DRAM realized by combining an open BL configuration and a folded BL configuration; FIGS. 3A through 3C are schematic diagrams illustrating the operation of the DRAM of FIG. 2; FIG. 4 is a schematic diagram of another conventional DRAM realized by combining an open BL configuration and a folded BL configuration; FIGS. 5A through 5C are schematic diagrams illustrating the operation of the DRAM of FIG. 4; FIGS. 6A through 6C are timing charts for driving the DRAM of FIG. 4; FIGS. 7A and 7B are schematic diagram of a conventional DRAM illustrating how inter-bit line noises are generated; FIG. 8 is a schematic circuit diagram of a conventional DRAM having an open/folded hybrid BL configuration; FIGS. 9A through 9C are diagrams showing different connections of bit lines, memory cells and sense amplifiers; FIG. 10 is a schematic circuit diagram of the surplus circuit used for a conventional DRAM; FIG. 11 is a graph of waveforms showing the operation of a conventional surplus circuit; FIG. 12 is a circuit diagram of a first embodiment of DRAM according to the invention; FIGS. 13A and 13B are timing charts for driving the first embodiment; FIG. 14 is a circuit diagram of a second embodiment of DRAM according to the invention; FIG. 15 is a circuit diagram of a third embodiment of DRAM according to the invention. FIGS. 16A and 16B are timing charts for driving the third embodiment; FIGS. 17A through 17D are circuit diagrams of exemplary drive circuits for generating control signals  0 through  4 to be used for the circuit of FIG. 15; FIGS. 18A through 18D are circuit diagrams of other exemplary drive circuits for generating control signals  0 through  4 to be used for the circuit of FIG. 15; FIG. 19 is a circuit diagram of still another exemplary drive circuit for generating control signals  0 through  4 to be used for the circuit of FIG. 15; FIG. 20 is a circuit diagram of a fourth embodiment of DRAM according to the invention; FIG. 21 is a timing chart for driving the fourth embodiment; FIG. 22 is a circuit diagram of a fifth embodiment of DRAM according to the invention; FIG. 23 is a circuit diagram of a sixth embodiment of DRAM according to the invention; FIG. 24 is a timing chart for driving the sixth embodiment; FIG. 25 is a circuit diagram of an exemplary drive circuit for generating control signals  0 through  4 to be used for the circuit of FIG. 23; FIGS. 26A and 26B are circuit diagrams of a seventh embodiment of DRAM according to the invention; FIG. 27 is a circuit diagram of an eighth embodiment of DRAM according to the invention; FIG. 28 is a circuit diagram of a ninth embodiment of DRAM according to the invention; FIG. 29 is a circuit diagram of a tenth embodiment of DRAM according to the invention; FIG. 30 is a circuit diagram of an eleventh reference example of a DRAM; FIG. 31 is a timing chart for driving the eleventh reference example; FIG. 32 is a circuit diagram of the eleventh reference example showing how the connections of the bit lines are switched; FIG. 33 is another circuit diagram of the eleventh reference example also showing how the connections of the bit lines are switched; FIG. 34 is still another circuit diagram of the eleventh reference example also showing how the connections of the bit lines are switched; FIG. 35 is still another circuit diagram of the eleventh reference example also showing how the connections of the bit lines are switched; FIG. 36 is still another circuit diagram of the eleventh reference example also showing how the connections of the bit lines are switched; FIG. 37 is still another circuit diagram of the eleventh reference example also showing how the connections of the bit lines are switched; FIG. 38 is a circuit diagram of a twelfth reference example of a DRAM; FIG. 39 is a circuit diagram of a thirteenth reference example of a DRAM. FIG. 40 is a timing chart for driving the thirteenth reference example; FIGS. 41A through 41D are circuit diagrams of exemplary drive circuits for generating control signals  0 through  4 to be used for the thirteenth reference example; FIG. 42 is a circuit diagram of another exemplary drive circuit for generating control signals  0 through  4 to be used for the thirteenth reference example; FIG. 43 is a circuit diagram of a fourteenth reference example of a DRAM; FIG. 44 is a schematic circuit diagram of the surplus circuit used for the fourteenth reference example; FIGS. 45A and 45B are circuit diagrams of circuit A and circuit B used for the surplus circuit of FIG. 44; FIG. 46 is a graph of waveforms showing the operation of the fourteenth reference example; FIG. 47 is a schematic circuit diagram of the surplus circuit used for a fifteenth reference example; FIGS. 48A through 48S are circuit diagrams of exemplary circuits that can be used for circuit C of the surplus circuit of FIG. 47; FIG. 49 is a circuit diagram of a sixteenth reference example of a DRAM; FIG. 50 is a schematic circuit diagram of the surplus circuit used for the sixteenth reference example; FIG. 51 is a circuit diagram of a seventeenth reference example of a DRAM; FIG. 52 is a schematic circuit diagram of the surplus circuit used for the seventeenth reference example; FIG. 53 is a schematic circuit diagram of the surplus circuit used for an eighteenth reference example; FIGS. 54A through 54C are circuit diagrams of exemplary barrel shifters that can be used for the surplus circuit of the eighteenth reference example; FIG. 55 is a schematic circuit diagram of the surplus circuit used for a nineteenth reference example; FIGS. 56A through 56D are circuit diagrams of exemplary barrel shifters that can be used for the surplus circuit of the nineteenth reference example; FIG. 57 is a schematic circuit diagram of the surplus circuit used for a twentieth reference example; FIG. 58 is a schematic circuit diagram of the surplus circuit used for a twenty first reference example; and FIGS. 59A and 59B are circuit diagrams of exemplary barrel shifters that can be used for the surplus circuit of the twenty first reference example. The present invention will be described in greater detail by referring to the accompanying drawings that illustrate preferred embodiments of the invention. (Embodiment 1) FIG. 12 is a circuit diagram of a first embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. FIGS. 13A and 13B are timing charts for driving the embodiment. This embodiment in fact provides an improvement to the DRAM illustrated in FIG. 2. The embodiment differs from the DRAM of FIG. 2 in that it additionally comprises control clocks  0 ,  1 ,  2 ,  3 ,  4 and  5 , of which  0 is connected to bit line /BLA of a corresponding sense amplifier section by way of capacitors C 1 and  1 is connected to bit line BL B of a corresponding sense amplifier section by way of capacitors C 1 . Similarly,  2 and  3 are respectively connected to bit lines /BL B and BL B of a corresponding sense amplifier section by way of capacitors C 2 , whereas  4 and  5 are respectively connected to bit lines /BL C and BL C of a corresponding sense amplifier section by way of capacitors C 1 , the capacitances C 1 and C 2 having different values (C 1 is not equal to C 2 ). Thus, assuming that WA 0 is selected to read out memory cell data onto BL 0 and BL 1 , the data on BL 0 is transferred to /BL A and sensed by SA 0 , using BL 2 as a reference bit line because P01 is at level "H". As the potential of BL 0 is raised because of the coupling of W A0 and BL 0 , occurring as a result of raising W A0 in addition to the cell data, SA 0 reads "1" with ease but "0" with difficulty. However, in this embodiment, since  1 is raised as shown in FIG. 13A, the capacitance of C 1 is coupled to the BL A side to raise the potential of BL A . Consequently, the situation where "1" is read with ease and "0" is read with difficulty is compensated to provide a same level of ease for reading both "1" and "0". Likewise, the data on BL 1 is transferred to BL B and sensed by SA 1 , using BL 5 as a reference bit line. As the potential of BL 1 is raised because of the coupling of W A0 and BL 1 , occurring as a result of raising W A0 in addition to the cell data, SA 1 reads "1" with ease but "0" with difficulty. However, since Ø2 is raised as shown in FIG. 13A, the capacitance of C 2 is coupled to the /BL A side to raise the potential of /BL B . Consequently, the situation where "1" is read with ease and "0" is read with difficulty is compensated to provide a same level of ease for reading both "1" and "0". Note that the level of ease with which SA 0 reads "1" differs from the level with which SA 1 reads "1". Similarly, the level of difficulty with which SA 0 reads "0" differs from the level with which SA 1 reads "0". For instance, both W A0 and BL 2 give rise to a small coupling when W A0 is raised (because of a parasitic capacitance existing at the intersection of BL 2 and W A0 although no memory cell is located there) so that the level of ease for SA 0 to read "1" is lower than the level of ease for SA 1 to read "1". This is due to the fact that reference bit line BL 2 FOR SA 0 is slightly raised because of the above described coupling. This difference can be eliminated by differentiating the capacitance value of C 1 from that of C 2 or selecting a capacitance value for C 2 smaller than that of C 1 . In additional, there may be a difference in the level of ease of reading "1" and the level of difficulty of reading "0" between SA 0 for folded reading and SA 1 for open reading attributable to the exposure to array noises and the difference in the worst reading pattern and such a difference can also be corrected by selecting values for C 1 and C 2 that satisfy C 1 =/ C 2 . FIG. 13B shows a timing chart with which the reference bit line is raised not by a coupling action but, conversely, by lowering the coupling of the bit line from which the data in the corresponding memory cell is read in order to correct the level of ease with which "1" is read and the level of difficulty with which "0" is read. In this case, a difference can also be corrected by selecting values for C 1 and C 2 that satisfy C 1 =/ C 2 . (Embodiment 2) FIG. 14 is a circuit diagram of a second embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. This embodiment differs from the first embodiment in that C 1 , C 2 , C 3 , C 4 , C 5 and C 6 in the sense amplifier sections have different capacitance values. As described above, the level of ease with which "1" is read and that of difficulty with which "0" is read by same SA 0 for folded reading may vary depending on which of word lines W A0 , W A1 and W A2 is selected. This situation can be appropriately corrected by selecting capacitance values that satisfy C 1 =/ C 2 , C 4 =/ C 4 and C 5 =/ C 6 . (Embodiment 3) FIG. 15 is a circuit diagram of a third embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. FIGS. 16A and 16B are timing charts for driving the embodiment. With this embodiment, the differences that may arise between the open reading and the folded reading and from the selection of one of W A0 through W A2 can be corrected by varying the amplitudes of control signals  0 through  5 , holding coupling capacitance C 1 of the sense amplifier sections. For instance, the difference between the reading of "1" and that of "0" varies for all the combinations of W A0 through W A2 of the open BL configuration and the folded BL configuration, this situation can be corrected by selecting values that satisfy V FA0 =/ V FA1 =/ V FA2 =/ V OA0 =/ V OA1 =/ V OA2 . As in the case of FIGS. 13A and 13B, the arrangement of FIG. 16A differs from that of FIG. 16B in that the difference is corrected by raising the reference side in the former while it is corrected by lowering the cell reading side. FIGS. 17A through 18D are circuit diagrams of exemplary drive circuits for generating control signals Ø0 through Ø3 to be used for the circuit of FIG. 15. Referring to FIGS. 17A through 17D,  FA0 goes to level "L" and the potential of  1 is shifted from V SS to V FA0 when W A0 is selected.  FA1 goes to level "L" and the potential of  0 is shifted from V SS to V FA1 when W A1 is selected. In this manner, the operation of the embodiment can be controlled by arranging reference potentials V FA0 through V FA2 and V OA0 through V OA2 . FIGS. 18A through 18D illustrates arrangements for dealing with a situation where the level of ease of reading "1" and that of reading "0" differ only on the open BL side and the folded BL side. Only two reference potentials V FA and V OA are required. FIG. 19 illustrates a circuit arrangement that requires no reference potential. With this circuit arrangement,  PRE is brought to "L" and capacitances C X2 and C X2 are precharged to V CC in the stand-by state. When, for example, W A1 is selected,  FA1 is brought to "L" to energize transistor Q 1 . If the capacitance of control signal is made equal to C Y under this condition, the potential of  0 shifts from V SS to C X2 /(C Y +C X2 )·V CC . As in the case of FIGS. 17A through 17D and FIGS. 18A through 18D, the amplitudes of the control signals can be varied freely by selecting C X2 =C X2 . Note that similar circuits can be arranged for  0 ,  2 and  3 . In a preliminary stage of operation,  PRE may be simply made equal to V CC and the discrepancy between the level of ease of reading "1" and that of reading "0" may be determined by externally applying a reference voltage to X 2 , X 2 to differentiate the values of C X2 and C X2 by that discrepancy so that the device may operate without external inputs for X 2 and X 2 . (Embodiment 4) FIG. 20 is a circuit diagram of a fourth embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. FIG. 21 is a timing charts for driving the embodiment. This embodiment is in fact provides an improvement to the DRAM illustrated in FIG. 4. Like the case of the first embodiment, this embodiment differs from the DRAM of FIG. 4 in that it additionally comprises control clocks  0 ,  1 ,  2 ,  3 ,  4 and  5 , which are connected to the respective bit lines of a corresponding sense amplifier section by way of capacitors C 1 and C 2 . As in the case of FIG. 12, the respective differences of the level of ease of reading "1" and that of reading "0" for SA 0 side (folded reading and folded rewriting) and those for SA 1 side (folded reading and open rewriting) can be corrected by selecting C 1 =C 2 . FIG. 21 shows an arrangement for slightly lowering the read-out side by coupling the sense amplifier section. (Embodiment 5) FIG. 22 is a circuit diagram of a fifth embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. If the levels of ease of reading "1" and "0" for W A0 , W A1 and W A2 differ from each other, they may be corrected by selecting C 1 =/ C 2 , C 3 =/ C 4 and C 5 =/ C 6 as in the case of FIG. 14. (Embodiment 6) FIG. 23 is a circuit diagram of a sixth embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. FIG. 24 is a timing charts for driving the embodiment. As in the case of the embodiment of FIG. 15, corrections can be made for this embodiment by varying the amplitudes of control signals  0 through  5 . FIG. 25 is a circuit diagram of an exemplary drive circuit for generating control signals  0 through  5 for the circuit of FIG. 23. Note that no reference potential is used for varying the amplitudes as in the case of FIG. 19. The potential of  0 of this embodiment is lowered by means of charge distribution from V CC . Note also that the circuit of FIG. 25 can also be used for drive operation of FIG. 16B. (Embodiment 7) FIGS. 26A and 26B are circuit diagrams of a seventh embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. With this embodiment, the level of ease of reading "1" and "0" can be corrected by arranging dummy cells in the cell arrays and selecting two or more than two values for lowering each of /DW A0 through /DW A2 and /DW B0 through /DW B2 or selecting three values for lowering one of /DW A0 through /DW A2 . Assume, for example, that BL 0 is raised more than BL 2 by VX as a result of various coupling operations when W A0 is selected. If BL 0 and BL 1 are lowered by V Y when dummy word line /DW A0 is lowered from V CC to V SS , then BL 1 and BL 2 will be lowered by V Y when /DW A1 is lowered from V CC to V SS . BL 0 and BL 2 will also be lowered by V Y when /DW A2 is lowered from V CC to V SS . Thus, if /DW A0 through /DW A2 are lowered by voltages V A0 /V CC , V A1 /V CC and V A2 /V CC , the difference between BL 0 and BL 2 after the correction of moving the dummy word lines will be expressed by the equation below. BL 0 - BL 2 = (V A0 x V Y + V A2 x V Y ) - (V A1 x V Y + V A2 × V Y ) = (V A0 - V A1 ) V Y On the open SA 1 side, the following equation holds true. BL 1 - BL 4 = (V A0 + V A1 ) V Y Thus, corrections can be carried out by selecting an appropriate value for V A0 + V A1 . (Embodiment 8) FIG. 27 is a circuit diagram of an eighth embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. With this embodiment again, the level of ease of reading "1" and "0" can be corrected by arranging dummy cells in the cell arrays and selecting two or more than two values for lowering each of /DW A0 through /DW A2 and /DW B0 through /DW B2 or selecting three values for lowering one of /DW A0 through /DW A2 . This embodiment operates essentially same as Embodiment 7 above. When W A0 is selected, the following equations hold true. BL 0 - BL 1 = (V A0 ' + V A1 ') V Y - (V A1 ' + V A2 ') V Y = (V A0 ' - V A2 ') V Y BL 0 - BL 2 = (V A0 ' + V A2 ') V Y - (V A1 ' + V A2 ') V Y = (V A0 ' - V A1 ') V Y So, the level of ease of reading "1" and that of reading "0" can be corrected appropriately. (Embodiment 9) FIG. 28 is a circuit diagram of a ninth embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. While capacitors are connected to the bit lines in the embodiment of FIG. 12, the same effect is achieved in this embodiment by modifying the gate length and the gate width of the switches arranged between the cell arrays and the sense amplifiers for the operation of the device. Since the switches are arranged to switch "H" and "L" for selected cells, the effect of the capacitors can be achieved simply by varying the degree of coupling without using additional clocks. (Embodiment 10) FIG. 29 is a circuit diagram of a tenth embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. While capacitors are connected to the bit lines in the embodiment of FIG. 20, the same effect is achieved in this embodiment by modifying the length and the width of the gate of the switches arranged between the cell arrays and the sense amplifiers for the operation of the device. (Reference Example 11) FIG. 30 is a circuit diagram of an eleventh reference example of a DRAM, showing two cell arrays (A, B) as examples. FIG. 31 shows timing charts for driving the circuit. Referring to FIG. 30, there are shown sense amplifiers SA 0 through SA 5 , word lines W A0 through W A2 and W B0 through W B2 , bit lines BL 0 through BL 15 arranged in cell arrays and bit lines /BL A0 , BL A0 , /BL A1 , BL A1 , /BL B0 , BL B0 , /BL B1 , BL B1 , /BL C0 , BL C0 , /BL C1 and BL C1 arranged in sense amplifier sections. There are also shown control signals P A0 , P A12 , P B11 , P B1 , P B020 , P B022 , P C11 , P C1 , P C020 , P C022 , P D0 and P D12 for switches designed to switch over the connections of the bit lines of memory cells and the bit line pairs of sense amplifiers. Sense amplifier (SA) with a folded BL configuration are arranged at the right end of cell array B and at the left end of cell array A. The SAs at the right end of cell array B are shared by the cell array located to the right of cell array B to form a shared SA system there. Likewise, the SAs at the left end of cell array A are shared by the cell array located to the left of the cell array A also to form a shared SA system there. As for the word lines of cell array A, assume now that W A0 is selected. Then, of SA 0 , SA 2 and SA 3 , SA 0 forms a pair of bit lines, using one of the three bit lines that has a memory cell, say BL 1 , and one of the bit line pair BL 3 and BL 5 of SA 3 having a folded BL configuration, say BL 3 located on the reference side, as folded BLs to operate the SA. Similarly, SA 2 forms a pair of bit lines of BL 2 and BL 0 as folded BLs to operate the SA and SA 3 forms a pair of bit lines of BL 5 and BL 3 as folded BLs to operate the SA. BL 3 is used as a reference bit line to be commonly used for SA 0 and SA 3 . For rewriting data into a memory cell, the shared state of the reference bit line is suspended and bit line pair BL 1 , BL 3 connected to sense amplifier SA 0 are used as folded BLs to carry out the rewriting operation, while one of the three bit lines of array B located next to BL 2 of array A, say BL 8 , is used as a reference bit line to be used for a bit line pair having an open BL configuration and connected to sense amplifier SA 2 . Likewise, one of the three bit lines of array B located next to BL 5 of array A, say BL 11 , is used as a reference bit line to be used for a bit line pair having an open BL configuration and connected to sense amplifier SA 3 . With the above arrangement, no noise that is specific to open BLs is generated by ways of word lines during reading operations to bring forth the effect of the folded BL configuration of canceling noises generated by way of unselected word lines to consequently reduce the overall noise level. Additionally, since the open BL configuration and the folded BL configuration are combined for rewriting data, it allows ordinary rewriting operations and equalization of bit lines to take place. Likewise, when a word line of array B is selected, a bit line of the array is used as a common reference for reading operation to effectuate a folded BL configuration. For a rewriting operation, on the other hand, one of the bit line pairs is used to effectuate a folded BL configuration, while the use of the other bit line pair as shared reference BLs is suspended and the bit line of array B through which the memory cell data is read out and the bit line (reference BL) of array A located adjacent thereto are used to form a pair of an open BL configuration for rewriting operation. The cell arrays have memory cells on two-third of the intersections of their word lines and bit lines so that, while the ratio of the number of memory cells to that of intersections is lower than that of cell arrays of an open BL configuration, it is higher than that of cell arrays of a folded BL configuration on which memory cells are arranged only on a half of the intersections of the word lines and the bit lines. Consequently, a memory device according to the invention can reduce the area of the memory cells if compared with a comparable memory device with a folded BL configuration. Additionally, a sense amplifier (SA) is arranged for every three bit lines (BL) to mollify the design requirements for SAs if compared with a device having an open BL configuration. Looking at a particular bit line pair of the arrangement FIG. 30 having a cell array arrangement as described above when it is used for reading data with the folded BL configuration, it will be understood that the bit line through which the data in a memory cell is read is placed between the bit line pair to change over the connection between the bit line of the memory cell and the bit line of the corresponding sense amplifier regardless of the location of the selected word line. For example, assume that word line W A0 is selected, it is raised after bringing P A12 , P B11 and P B1 to level "L" and keeping P A0 , P B020 and P B022 to level "H" as shown in FIG. 31. Then, the memory cell data are read onto BL 1 , BL 2 , BL 3 and BL 4 . Then, data are taken into SA 0 by using BL 1 through which the corresponding cell data is read out and reference bit line BL 3 as a folded BL pair because P A0 , P B020 and P B022 are held to "H". Similarly, data are taken into SA 1 by using BL 4 through which the corresponding cell data is read out and reference bit line BL 6 as a folded BL pair and also taken into SA 2 by using BL 5 through which cell data is read out and reference bit line BL 3 as a folded BL pair. Note that reference bit line BL 3 is shared by the bit line pairs of SA 0 and SA 3 . While only four columns and hence only one shared reference bit line are shown in FIG. 30, it will be understood that a shared reference bit line appears for every three bit lines. This arrangement apparently differs from that of FIG. 4 since, in this arrangement, bit line BL 2 through which the cell data of SA 2 is read out is placed between bit line pair BL 1 , BL 3 of SA 0 and bit line BL 5 through which the cell data of SA 3 is placed between bit line pair BL 4 , BL 6 of SA 3 while bit line BL 1 through which the cell data of SA 0 is read out is placed between bit line pair BL 0 , BL 2 of SA 2 and bit line BL 4 through which the cell data of SA 1 is read out is placed between bit line pair BL 3 , BL 5 . The above described data reading operation that takes place when word line W A0 is selected may be understood more clearly by referring to emphatically drawn solid lines of FIG. 32. As BL 2 through which the memory cell data that is put into SA 2 is read out is subjected to a change equal to the volume of the read signal to give rise to a noise that is applied to the adjacent bit lines by way of the coupling capacitance C BB between the bit lines (FIG. 32). Now, since the bit line pair BL 1 and BL 3 of SA 0 are located on the both sides of BL 2 , they are subjected to an equal amount of noise with a same phase. If seen from SA 0 , the potential difference between BL 1 and BL 3 that form a bit line pair does not change and the effect of the array noise of BL 2 is canceled. Thus, the net noise that bit line pair of BL 1 , BL 3 is subjected to is the inter-bit line noise of C BB V S (V S representing the volume of the read signal of BL 4 ) generated by BL 4 through which the memory cell data is read out for SA 1 . In other words, the net noise can be reduced to one-third of the inter-bit line noise of 3C BB V S of FIG. 4 and, because the noise level can be lowered without using twisted bit lines, the overall chip size can be significantly reduced. For the same token, the inter-bit line noise attributable to the read out signals from the bit lines placed between the bit line pairs of SA 1 , SA 2 and SA can be canceled so that the inter-bit line noise of C BB V S on any of the bit lines can be reduced to one-third of the inter-bit line noise of 3C BB V S of FIG. 4. After the date of a memory cell is read out onto a corresponding sense amplifier, P A0 , P B020 and P B022 are brought to level "L" as shown in FIG. 31 to activate the sense amplifier to amplify a minute signal. Now, let us look into the operation of putting back or rewriting the data onto the original memory cell. P A0 is brought to level "H" for SA 0 and SA 1 and data are rewritten into the corresponding memory cells by using BL 1 and BL 3 and BL 4 and BL6 as bit line pairs with a folded BL configuration. Under this condition, since BL 3 is used for SA 0 and cannot be used for the rewriting operation of SA 3 , P B020 is brought back to level "H", P B022 is held to level "L" and P C022 connected to adjacent array B is brought to level "H" for the rewriting operation on SA 2 and SA 3 . In other words, SA 2 uses BL 2 and BL8 for a bit line pair with an open BL configuration and SA 3 uses BL 5 and BL 11 for a bit line pair also with an open BL configuration. Likewise, when word lines W A1 and W A2 are selected, control signals P A0 , P A12 , P B1 , P B11 , P B020 , P B022 , P C11 , P C1 , P C020 and P C022 are controlled appropriately as shown in FIG. 31 so that, when W A1 is selected, the connection between the bit line for reading the data of the memory cell and the bit line of the corresponding sense amplifier is switched to that of FIG. 34, while it is switched to that of FIG. 35 for a rewriting operation. Similarly, when W A2 is selected, the connection between the bit line for reading the data of the memory cell and the bit line of the corresponding sense amplifier is switched to that of FIG. 36, while it is switched to that of FIG. 35 for a rewriting operation. In either case, a bit line through which data is read out of a memory cell is switched and placed between a bit line pair for every data reading operation to reduce the maximum array noise may be always reduced to C BB V S . In FIG. 36, BL 1 and BL6 are respectively placed between bit line pair BL 0 and BL 2 of the outermost sense amplifier SA 2 and bit line pair BL 5 and BL 7 of sense amplifier SA 1 These bit lines are not used for reading out memory cell data onto any sense amplifier when W A2 is selected and hence do not affect bit line pairs of BL 0 and BL 2 and BL 5 and BL7 with reading out noises. It may be understood by comparing FIGS 32, 34 and 36 that, by switching the word line from W A0 to W A1 , the selected memory cell moves downward by a bit line in the memory cell and, correspondingly, the connection of the bit line of the memory cell array and that of the sense amplifier is also switched downward by a bit line when W A1 is selected. When W A2 is selected, however, the connection of bit lines is lowered by a bit line on the side of SA 0 and SA 1 , although it is made same as when W A0 is selected on the side of SA 2 and SA 3 . With this arrangement of switching the connection of the bit line of the cell array and that of the sense amplifier according to the selected word line, the noise level can be significantly reduced. (Reference Example 12) FIG. 38 is a circuit diagram of a twelfth embodiment of DRAM according to the invention, showing two cell arrays (A, B) as examples. This embodiment is the same as the arrangement shown in FIG. 30 in terms of equivalent circuit and differs from it in that the switches for connecting the sense amplifiers and the memory cells show a mirror image of those of FIG. 30 in the sense of word lines within the broken circles. In other words, the switches may be located differently without changing the equivalent circuit. Since the operation of the circuit of FIG. 38 operates exactly same as that of the circuit of FIG. 30, it will not be described here any further. (Reference Example 13) FIG. 39 is a circuit diagram of the thirteenth reference example of DRAM, showing two cell arrays (A, B) as examples. FIG. 40 is a timing chart for driving the circuit. This circuit is free from the following problem of the eleventh reference example. By looking into the operation of SA 0 when word line W A0 is selected in FIG. 32, it will be understood that P A12 , P B11 and P B1 are brought down from "H" to "L" before the word line is raised for reading data. Under this condition, the bit line connected to the source or drain is slightly lowered by capacitive coupling of the gate and the source or the gate and the drain of the transistor of the switch between the cell array and the sense amplifier. This is indicated by "o" in FIG. 32. Additionally, since the cell data is latched to SA 0 after the word line is raised arid it is moved into SA 0 , P A0 , P B020 and P B022 are lowered. The locations to be affected by coupling are indicated by "x". Note that bit line /BL A0 through which the memory cell data of SA 0 is read out carries three "o"s and a "x" and reference bit line BL A0 also carries three "o"s and a "x". On the other hand, when W A1 is selected as shown in FIG. 34, bit line /BL A0 on the cell data side of SA 0 carries two "o"s and a "x", while bit line BL A0 on the reference side carries three "o"s and a "x". When W A2 is selected as shown in FIG. 36, bit line BL A0 on the cell data side of SA 0 carries three "o"s and a "x", while bit line /BL A0 on the reference side carries two "o"s and a "x". Thus, the influence of coupling differs depending on the selected one of bit lines W A0 through W A2 and consequently the level of ease of reading data "1" differs from that of reading data "0". For instance, when a bit line through which memory cell data is read out is lowered as it is influenced by coupling, "1" becomes less easy to be read, whereas "0" becomes easier to be read. Additionally, the level of ease of reading data "1" may differ from that of reading data "0" when some of the components such as switches are arranged differently as in the case of FIGS. 30 and 38. The arrangement of FIG. 39 can dissolve such problems. In FIG. 39, capacitors having a coupling capacitance of C 1 are connected to bit line pairs of the sense amplifier sections and control signals  0 through  5 are connected to the opposite terminals of the respective capacitors so that the above described unbalance of bit line pairs is compensated by using  0 through  5 when data are read out from memory cells. It will be understood that the idea is the same as that of the first embodiment described earlier. Since the degree of unbalance of bit line pairs may vary depending on the selected one of word lines W A0 through W A2 , the amplitudes of ⊘0 through ⊘5 are differentiated depending on the selected one of word lines W A0 through W A2 as shown in FIG. 40. Thus, the technique used in this arrangement differs from any known dummy cell techniques or coupling techniques of a single correction type with which all the amplitudes of the control signals are made same and identical. Referring to FIG. 40, as the data of a memory cell is read out, the control signal connected to C 1 of the bit line of the sense amplifier section located on the bit line side of the memory cell from which the data is read out is lowered. For example, when W A0 is selected, the memory cell data is read out onto /BL A0 so that  0 is lowered to V FFA0 . FIGS. 41A through 41D illustrate exemplary circuits that can realize such a variable amplitude. Signal  0 is precharged to V CC because  FFA0 and  FFA1 are at level "L". When W A0 is selected, only V FFA0 is raised to level "H" and  0 is lowered to the source voltage of V FFA0 . When different source voltages of V FFA0 , V FFA1 , V FFA2 , V FOA0 , V FOA1 and V FOA2 are generated in the chip in FIGS. 41A through 41D, the circuit of FIG. 42 can generate any desired voltage from V CC and V SS in a simple manner. Referring to FIG. 42, at the time of precharging, / PRE is at "H" and C X0 and C X2 are precharged to V SS , while  is precharged V CC because  FFA0 and  FFA1 are at level "L". Then, as / PRE goes to level "L" and  FFA0 rises to level "H" as a result of the selection of W A0 , output  0 falls as a result of charge distribution of capacitance C X0 and negative capacitance C Y . The amplitude of  0 can be varied freely by changing the values of C X0 and C X2 . This technique can be applied to the arrangement of FIG. 38. It should be noted that, not only the amplitudes of  0 through  2 can be independently varied but also the location of the word line used to select the value of capacitance C 1 of the sense amplifier, the locations of the sense amplifiers for receiving  0 and  1 , capacitance C 1 of the sense amplifiers for receiving  2 ,  3 ,  4 and  5 , the amplitude of dummy word lines and the potentials of dummy cells if they are of the type of external data writing in order to correct any undesired unbalances. The following examples are designed to improve the logic circuit for determining the remainder of the division of the address of a word line by three, the output of which logic circuit is used as a signal for driving memory cell arrays for the purpose of the invention. Note that such a logic circuit may also be used for Embodiments 1 through 13 described above. (Reference Example 14) FIG. 43 is a circuit diagram of a fourteenth reference example of a DRAM. This arrangement is a dynamic semiconductor memory device with an open/folded hybrid BL configuration, wherein each of the cell arrays comprises a total of 256 word lines. The basic configuration of the device is not particularly different from that of any comparable conventional devices. A 8-bit address of A 0 through A 7 is used for the word lines. FIG. 44 is a circuit diagram of a surplus circuit for determining the remainder produced by dividing the address of the selected word line by three. In this arrangement, three signal lines Z 0 , Z 1 and Z 2 are used for the output of the surplus circuit as in the case of the known surplus circuit described earlier. More specifically, Z 0 ="1" is used if the remainder of the division by three of the address is equal to 0 and Z 1 ="1" is used if the remainder of the division by three of the address is equal to 1, whereas Z 2 ="1" is used if the remainder of the division by three of the address is equal to 2. In addition, that the chip is in a stand-by state is expressed by Z 0 =Z 1 =Z 2 ="0". The relationship between the output of the surplus circuit and the input address is shown in Table 2 below. A 7 A 6 A 5 A 4 A 3 A 2 A 1 A 0 Z 0 Z 1 Z 2 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 0 0 0 1 0 1 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 1 0 1 0 0 1 0 0 0 0 0 1 1 0 1 0 0 0 0 0 0 0 1 1 1 0 1 0 0 0 0 0 1 0 0 0 0 0 1 ... ... 1 1 1 1 1 1 1 1 0 1 0 0 0 0 when chip is in a stand-by state The surplus circuit comprises circuits A and B. The eight bits of each address is divided into four 2-bits, each of which is divided by three and the remainder is determined by circuit A, whereas circuit B adds two 3-phase signals and divides the sum by three to produce the remainder. FIGS. 45A and 45B shows exemplary circuit configurations of circuit A and circuit B respectively. Address signal A 5 and its complementary signal /A i are used as inputs to the surplus circuit. These address signals are held to "L" when the chip is in a stand-by state and either A i or /A i goes to "H" when the chip is activated. When the chip is in a stand-by state and hence A i and /A i are held to "L", any outputs from circuit A are at "L" to make any outputs from circuit B that uses the outputs of circuit A are also at "L" to consequently make all the outputs of Z 0 , Z 1 and Z 2 of the surplus circuit stay at "L". When the chip is activated and either A i or /A i is brought up to "H", only one of the three outputs of circuit A goes to "H". Then, the result is conveyed to circuit B and, in the end, only one of the three outputs of the surplus circuit changes from "L" to "H". FIG. 46 illustrates the above description. Since Z 0 , Z 1 and Z 2 are all at "L" in a stand-by state, no problem of multiple selection as illustrated in FIG. 11 occurs when the chip is activated so that the chip is made to operate stably at any time. (Reference Example 15) FIG. 47 illustrates a fifteenth reference example , which is a surplus circuit having circuit A that uses 4-bit addresses as inputs. FIGS. 48A through 48S illustrates exemplary circuit diagrams of circuits C of the surplus circuit of FIG. 47. The surplus circuit comprises circuits C for receiving a 4-bit address signal and its complementary signal and produces the remainders obtained by dividing them by three respectively and circuit B for receiving the outputs of two circuits C and produces the remainder obtained by dividing the sum by three. Note that circuit B is identical with the one used in the fourteenth reference example. In this reference example, three Y 00 , Y 01 and Y 02 of the first circuit C that receives address signals A 0 through A 3 and three outputs Y 10 , Y 11 and Y 12 of the second circuit C that receives their complementary signals A 4 through A 7 are applied to circuit B that is identical with that of the fourteenth reference example and outputs Z 0 , Z 1 and Z 2 of circuit B represents the remainders obtained by dividing addresses A 0 through A 7 by three. As in the case of the preceding example, when the chip is in a stand-by state and address signal A i and its complementary signal /A i is held to "L", any outputs from circuit C is held to "L" so that, consequently, all the outputs Z 0 , Z 1 and Z 2 of circuit C is held to "L". The relationship between the input address and the output of the surplus circuit is the same as that of the preceding example. Note that each address is divided into 2-bits in the fourteenth example and the remainders obtained by dividing them by three are added by the addition circuit of circuit B, whereas each address is divided into 4-bits, which are then added in the fifteenth example. Likewise, addresses may be divided into even-bits such as 2-bits, 4-bits, 6-bits and so on for determining the remainders of the division by 3 of addresses. On the other hand, the maximum number of addresses from which remainders are obtained is typically the number of the word lines comprises in a memory cell array. In other words, the number of bits of addresses is typically eight (256 word lines) to ten (1,024 word lines). Thus, an address is preferably divided into 2-bits or 4-bits. (Reference Example 16) FIG. 49 is a circuit diagram of a sixteenth reference example of a DRAM. In this embodiment, each memory cell array comprises 512 word lines, although it is a dynamic semiconductor memory device with an open/folded hybrid BL configuration like Examples 14 and 15. Addresses A 0 through A 8 of the word lines are 9-bit addresses. FIG. 50 shows a circuit diagram of the surplus circuit for determining the remainder obtained by dividing the address of a selected word line by three. The relationship between the output of the surplus circuit and the input address is shown in Table 3 below. A 8 A 7 A 6 A 5 A 4 A 3 A 2 A 1 A 0 Z 0 Z 1 Z 2 0 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 0 0 0 0 1 0 1 0 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 0 1 1 1 0 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 0 1 0 1 0 0 1 0 0 0 0 0 0 1 1 0 1 0 0 0 0 0 0 0 0 1 1 1 0 1 0 0 0 0 0 0 1 0 0 0 0 0 1 ··· ··· 1 1 1 1 1 1 1 1 1 0 0 1 when chip is in a stand-by state 0 0 0 (Reference Example 17) FIG. 51 is a circuit diagram of a seventeenth reference example of a DRAM. In this circuit, each memory cell array comprises 1,024 word lines, although it is a dynamic semiconductor memory device with an open/folded hybrid BL configuration like Examples 14 through 16. Addresses A 0 through A 9 of the word lines are 10-bit addresses. FIG. 52 shows a circuit diagram of the surplus circuit for determining the remainder obtained by dividing the address of a selected word line by three. The relationship between the output of the surplus circuit and the input address is shown in Table 4 below. A 9 A 8 A 7 A 6 A 5 A 4 A 3 A 2 A 1 A 0 Z 0 Z 1 Z 2 0 0 0 0 0 0 0 0 0 0 1 0 0 0 0 0 0 0 0 0 0 0 1 1 0 1 0 0 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 0 0 1 1 1 1 0 0 0 0 0 0 0 0 0 1 0 0 0 1 0 0 0 0 0 0 0 0 1 0 1 0 0 1 0 0 0 0 0 0 0 1 1 0 1 0 0 0 0 0 0 0 0 0 1 1 1 0 1 0 0 0 0 0 0 0 1 0 0 0 0 0 1 ... ... 1 1 1 1 1 1 1 1 1 1 1 0 1 0 when chip is in a stand-by state 0 0 0 (Reference Example 18) FIGS. 53 and 54A through 54C illustrate an eighteenth peference example which is a surplus circuit comprising barrel shifters utilizing a pass transistor logic for switching first three inputs X 0 , X 2 and X 2 and three outputs Y 0 , Y 1 and Y 2 depending on the value of a second input. FIG. 53 is a circuit diagram illustrating the connection of barrel shifters and a precharge circuit connected to the output nodes of the barrel shifters, whereas FIGS. 54A through 54C illustrate individual barrel shifters respectively. The barrel shifters of FIGS. 54A through 54C control the connection of first inputs X 0 , X 2 and X 2 and outputs Y 0 , Y 1 and Y 2 by means of second inputs of 2-bit address signals A i and A j and their complementary signals /A i and /A j . In this embodiment, X 0 , X 2 and X 2 and Y 0 , Y 1 and Y 2 are connected by means of nMOS and pMOS transistors. The nMOS transistor has a gate length of LN and a gate width of WN, whereas pMOS transistor has a gate length of LP and a gate width of WP. While a CMOS comprising an nMOS and a pMOS is effective for transmitting all input voltages without discrimination using a threshold value, the same effect can be achieved to produce similar outputs simply by means of either an nMos or a pMOS. The relationship between inputs X 0 , X 2 and X 2 and outputs Y 0 , Y 1 and Y 2 is shown in Table 4 below. Y 0 Y 1 Y 2 A i =0, A j =0 (0) X 0 X 2 X 2 A i =0, A j =1 (1) X 2 X 0 X 2 A i =1, A j =0 (1) X 2 X 2 X 0 A i =1, A j =1 (3) X 0 X 2 X 2 When the chip is in a stand-by state, all address signals A i , A j , /A i and /A j are at "L" so that the first input and the output of circuit A are electrically isolated from each other. Since the output of the fourth barrel shifter in FIG. 53 is brought into an electrically floating state under this condition, the potentials of output Z 0 , Z 1 and Z 2 are held stationary by means of a precharge circuit. More specifically, since address signal A 0 and its complementary signal /A 0 is held to "L" in a stand-by state, outputs Z 0 , Z 1 and Z 2 are short-circuited to show the ground potential under this condition. When the chip is activated, either A 0 or /A 0 is brought up to "H" and, therefore, Z 0 , Z 1 and Z 2 are isolated from the ground by the precharge circuit. However, their output nodes are connected to the input potentials by the barrel shifter connected thereto in series under the activated condition, the output potential would not become unstable. The relationship between address signals A 0 through A 7 and outputs Z 0 , Z 1 and Z 2 of the surplus circuit is shown in Table 6 below. A 7 A 6 A 5 A 4 A 3 A 2 A 1 A 0 Z 0 Z 1 Z 2 0 0 0 0 0 0 0 0 H L L 0 0 0 0 0 0 0 1 L H L 0 0 0 0 0 0 1 0 L L H 0 0 0 0 0 0 1 1 H L L 0 0 0 0 0 1 0 0 L H L 0 0 0 0 0 1 0 1 L L H 0 0 0 0 0 1 1 0 H L L 0 0 0 0 0 1 1 1 L H L 0 0 0 0 1 0 0 0 L L H ... ... 1 1 1 1 1 1 1 1 L H L L L L when the chip is in a stand-by state (Reference Example 19) FIGS. 55 and 56A through 56D shows a nineteenth reference example. FIG. 55 shows the connection of barrel shifters and FIGS. 56A through 56D are circuit diagrams of the individual barrel shifters. In this arrangement each of the barrel shifters is provided with a precharge circuit for holding the output voltage to a fixed level in a stand-by state. More specifically, when all input signals A i , A j , /A i and /A j are at "L", X 0 , X 2 and X 2 and Y 0 , Y 1 and Y 2 are electrically insulated from each other and only the outputs are held to the ground potential. The relationship between the addresses and the outputs is exactly same as those of examples 14 through 16. Note that, with circuit configuration of this embodiment, the gate length and the gate width of each of the barrel shifters that are connected in series do note necessarily be held constant. The ratio of gate width W to gate length L, or β=W/L, is normally proportional to the drive capacity of transistor so that the delay time from the input to the output of the surplus circuit can be reduced by varying the ratios of the four barrel shifters connected in series slightly for each of them. More specifically, the delay time can be minimized by selecting β values of the pMOS transistors of the four barrel shifters A, B, C and D that are connected in series so as to satisfy the following relationship. β PA > β PB > β PC > β PD This is because each of the barrel shifters has its own electric capacitance and, therefore, the electric charge it has to allow to pass therethrough for electric charge from "L" in the precharge stage to "H" for a selected condition increases with the distances from outputs Z 0 , Z 1 and Z 2 . Note that the operating speed of nMOS transistor is not related to this embodiment in terms of β. So, the following relationship may well be satisfied. β NA = β NB = β NC = β ND (Reference Example 20) FIG. 57 shows a twenties reference example. In this arrangement, the outputs of the circuit A of example 14 are applied as input to the barrel shifters of example 18. The relationship between addresses and outputs is exactly same as those of examples 14 through 17. (Reference Example 21) FIGS. 58, 59A and 59B show a twenty first reference example. Unlike examples 18 through 20, where 2-bit addresses are applied to a barrel shifter, it has barrel shifters to show a one-to-one correspondence with addresses. Note, however, two types of shifters are provided. A first shifter S1 shifts the output by +1 relative to the input for address A i ="1" (Y 2 =X 2 , Y 1 =X 0 , Y 0 =X 2 ) but does not shift the output relative to the input for address A i ="0" (Y 2 =X 2 , Y 1 =X 2 , Y 0 =X 0 ) and a second shifter S2 shifts the output by -1 relative to the input for address A i ="1" (Y 2 =X 0 , Y 1 =X 2 , Y 0 =X 2 ) but does not shift the output relative to the input when address A i ="0". The first shifter S1 is used for even number addresses (A 0 , A 2 , A 4 , A 6 ), whereas the second shifter S2 is used for odd number addresses (A 1 , A 3 , A 5 , A 7 ). FIGS. 59A and 59B are circuit diagrams of the shifters S1 and S2. As in the case of Example, 18, a precharge circuit is provided at the final stage (Z 0 , Z 1 , Z 2 ) of output of the surplus circuit. When the chip is in a stand-by state, the relationship of Z 2 =Z 1 =Z 0 ="L" is secured. However, note that the single precharge circuit may be replaced by individual precharge circuits as in the case of Example 19.

Claims: A dynamic semiconductor memory device comprising: at least first (A) and second (B) memory cell arrays having a plurality of memory cells arranged at respective intersections of a plurality of word lines (WL) and a plurality of bit lines (BL); a first sense amplifier section (SA0) connected at an end of the first cell array to a plurality of first bit line pairs formed by part of said plurality of bit lines of said first cell array, said plurality of first bit line pairs having a folded bit line configuration; and a second sense amplifier section (SA1) connected to a plurality of second bit line pairs, each of which is formed by one bit line of the remaining bit lines of said first cell array and one bit line of part of the plurality of bit lines of said second cell array, said plurality of second bit line pairs having an open bit line configuration; characterised in that said memory device further comprises shifting means (,C) adapted to correct imbalance between the read voltages of bit line pairs in an array, said shifting means (,C) comprising a plurality of control signal (Φ) lines and a plurality of capacitors (C), wherein each one of the plurality of capacitors (C) is associated with a respective bit line of the plurality of bit lines (BL), each one of the plurality of capacitors (C) having a terminal connected to its associated bit line and another terminal connected to one of the plurality of control signal (Φ) lines, said shifting means shifting the read voltage on bit lines on which binary data is read out or on reference bit lines. A dynamic semiconductor memory device comprising: at least first (A) and second (B) memory cell arrays having a plurality of memory cells arranged at respective intersections of a plurality of word lines (WL) and a plurality of bit lines (BL); and a sense amplifier section (SA) connected to an end of the first cell array to a plurality of bit line pairs formed by part of said plurality of bit lines of said first cell array, wherein said plurality of bit line pairs have a folded bit line configuration, and the remaining bit lines of said first cell array form bit line pairs respectively with the reference bit lines of said bit line pairs having a folded bit line configuration to show a folded bit line configuration until the sense amplifiers start operating and then form bit line pairs having open bit line configuration respectively with part of the plurality of bit lines of a second cell array located adjacent to said first cell array with a sense amplifier section interposed between first and second memory cells arrays during a subsequent restoring operation which rewrites data into the memory cells after the operation of the sense amplifiers, characterised in that said memory device further comprises shifting means (,C) adapted to correct imbalance between the read voltages of bit line pairs in an array, said shifting means (,C) comprising a plurality of control signal (Φ) lines and a plurality of capacitors (C), wherein each one of the plurality of capacitors (C) is associated with a respective bit line of the plurality of bit lines (BL), each one of the plurality of capacitors (C) having a terminal connected to its associated bit line and another terminal connected to one of the plurality of control signal (Φ) lines, said shifting means shifting the read voltages on bit lines on which binary data is read out or on reference bit lines. A dynamic semiconductor memory device according to claim 1 or 2, characterised in that the bit lines having the folded bit line configuration and those having the open bit line configuration change connection to said sense amplifiers to take a predetermined bit line configuration according to the positions of the selected word lines. A dynamic semiconductor memory device according to claim 1 or 2, characterised in that two-thirds of the bit lines have a folded bit line configuration and the remaining one-third have an open bit line configuration. A dynamic semiconductor memory device according to claim I or 2, characterised in that a bit line with an open bit line configuration is placed between each bit line pair with a folded bit line configuration. A dynamic semiconductor memory device according to claim 1 or 2, characterised in that a set of lines is formed by every three bit lines, two of which form a bit line pair with a folded bit line configuration and the remaining bit line form an open bit line pair with a bit line of an adjacent cell array. A dynamic semiconductor memory device according to claim 1 or 2, further comprising shifting means for correcting each of the voltage differences between "1" data read out voltage and reference bit line voltage and between "0" data read out voltage and reference bit line voltage if there are more than one such differences. A dynamic semiconductor memory device according to claim 1 or 2, characterised in that said shifting means performs at least either a capacitor coupling operation or an electric charge distributing operation for either the bit line or the reference bit line of a memory cell selected for reading data therefrom. A dynamic semiconductor memory device according to claim 1 or 2, wherein said shifting means have more than two levels of ease of reading data. A dynamic semiconductor memory device according to claim 1, wherein correction amounts corrected by said shifting means are different between an open bit line structure and a folded bit line structure. A dynamic semiconductor or memory device according to claim 1 or 2, wherein correction amounts corrected by said shifting means are different in types of selected word lines. A dynamic semiconductor or memory device according to claim 1 or 2, wherein two-thirds of the intersections of the plurality of word lines and the plurality of bit lines carry respective memory cells. A dynamic semiconductor memory device according to claim 5, further comprising a switch (P) for changing over the connection between the bit line in the memory cell and said sense amplifier in order that said bit line with an open bit line configuration is placed between said each bit line pair with a folded bit line configuration.

Inventors: INABA TSUNEO, OHSAWA TAKASHI, OOWAKI YUKIHITO, SHIRATAKE SHINICHIRO, TAKASHIMA DAISABURO, INABA, TSUNEO, OHSAWA, TAKASHI, OOWAKI, YUKIHITO, SHIRATAKE, SHINICHIRO, TAKASHIMA, DAISABURO, Inaba, Tsuneo, c/o K.K. Toshiba, Ohsawa, Takashi, c/o K.K. Toshiba, Oowaki, Yukihito, c/o K.K. Toshiba, Shiratake, Shinichiro, c/o K.K. Toshiba, Takashima, Daisaburo, c/o K.K. Toshiba
Applicants: TOSHIBA KK, KABUSHIKI KAISHA TOSHIBA
Application year: 1996
Main classification: G11C11
IPC codes: G11C11, H01L21, H01L27