Biquadratic basic cell for programmable analog time-continuous filter

Abstract: The invention relates to an elementary biquadratic cell for programmable time-continous analog filters, which is placed between a first supply voltage reference (Vdd) and a second voltage reference (GND) and is of a type having at least one pair of input terminals (I31,I31') and first (O31,O31') and second (O32,O32') pairs of output terminals, and having a pair of half-cells (31,31'), which half-cells are structurally identical with each other and each comprised of at least a first transistor (T31,T31') placed between the first (Vdd) and the second (GND) voltage reference and having a base terminal connected to a respective one of the input terminals (I31,I31'). Each half-cell (31,31') further comprises second (T32,T32') and third (T33,T33') transistors placed between the first (Vdd) and second (GND) voltage references, the second transistor (T32,T32') having a base terminal connected to the first output terminal (O31,O31') of the first pair of output terminals and a collector terminal connected to the second output terminal (O32,O32') of the second pair of output terminals, and the third transistor (T33,T33') having an emitter terminal connected to the first output terminal (O31,O31') of the first pair of output terminals ad a base terminal connected to the second output terminal (O32',O32) of the second pair of output terminals of the other half-cell.

Description: Field of the Invention This invention relates to an elementary biquadratic cell for programmable time-continous analog filters. In particular, the invention concerns a biquadratic cell which is placed between a first supply voltage reference and a second voltage reference and has at least one pair of input terminals and first and second pairs of output terminals, said cell being formed by a pair of half-cells which are structurally identical, each comprising at least a first transistor placed between the first and the second voltage reference and having a base terminal connected to a respective one of the input terminals. Background Art Reference will be made hereinafter in particular, but not exclusively, to signal analog filters for applications related to the reading phase of data from mass memories (also referred to in the technical literature as 'disk-drive applications'). In this field, the filter cutoff frequencies referred to herein are currently on the order of 50 to 100 MHz. As is well known, the provision of integrated electronic filters in high-frequency applications bring some problems in using filtering techniques which employ configurations of feedback operational amplifiers intended to provide a desired transfer function. In fact, the frequency limitations of operational amplifiers appear as parasitic single points in the filter transfer function, and this limits the filter performance. This, first problem can be circumvented at certain frequencies by using open ring configurations, that is to say: using filters formed by transconductors and capacitors, referred to as gm/C's for brevity. Most analog filters can be formed by elementary cells of the low-pass type. For example, Figure 1 shows a biquadratic cell 10 according to this prior art. The cell 10 has first 1, second 2, third 3 and fourth 4 differential transconductors, connected in parallel together between a pair of input terminals I1, I2 and a pair of output terminals O1, O2 of the biquadratic cell 10. Each differential transconductor, 1, 2, 3 and 4, has a respective transconductance value gm1, gm2, gm3 and gm4, and has first I+ and second I- input terminals and first O+ and second O- output terminals. The first 1 and second 2 differential transconductors have their output terminals connected to a voltage reference, specifically a signal ground GND, through a first C1, C1' and a second C2, C2' pair of capacitors, respectively. Also, the third 3 and fourth 4 differential transconductors have their output terminals feedback connected to the output terminals of the second 2 and the first 1 differential transconductor, respectively. From a functional standpoint, the fourth differential transconductor, and the first capacitor pair C1, C1', operate as a pure integrator, whereas the second 2 and third 3 differential transconductors, and the second capacitance pair C2, C2' operate as a damped integrator. It should be noted that the third differential transconductor 3, being closed into a unitary feedback, is equivalent to a resistor having a value of 1/gm3. In addition, the first differential transconductor 1 provides the needed voltage-to-current conversion at the input of the biquadratic cell 10. Each differential transconductor 1, 2, 3 and 4 is implemented as a symmetrical differential structure, shown in Figure 2 in connection with the state of the art. In particular, a differential transconductor is shown at 1 in Figure 2, for example, which is placed between a supply voltage reference Vdd and the ground GND. The differential transconductor 1 has a pair of input terminals I+, I- and a pair of output terminals O+. O-, its structure being symmetrical with respect to an internal node X. The differential transconductor 1 includes, in fact, respective bias current generators G1, G2 which are placed between the supply voltage reference Vdd and respective bipolar transistors T1, T2, the latter having their base terminals connected to the input terminals I+, I- and their collector terminals connected to the output terminals O+, O-. The differential transconductor 1 further includes respective resistors R1, R2 of same value R, which are placed between the emitter terminals of the bipolar transistors T1, T2 and the internal node X, the latter being connected to the ground GND through a common bias current generator GC. In particular, these resistors R1, R2 may advantageously be MOS transistors working in their linear zone. The differential structure of Figure 2 allows the equivalent transconductance value Gmeq of the differential transconductor 1 to be varied by changing the value R of the resistors R1 and R2; the Gmeq value being given by: Gmeq = gmbip/(1+gmbip*R) where, R is the value of the resistors R1 and R2; and gmbip is the transconductance value of the bipolar transistors T1 and T2. Thus, by using resistors R1, R2 formed of MOS transistors, the value R can be varied in a simple manner by changing a tuning current for such MOS transistors. It matters to observe that the current draw of a biquadratic cell is the same as that of four differential transconductors, which transconductors must be dimensioned to suit the linear performance required for a given dynamic range of the signals to be processed. In fact, since the behavior of the transconductor stages cannot be linearized by the provision of appropriate feedback loops, the input signal is only required to modulate a small fraction of the current at rest through the active components, including essentially the bipolar transistors. For example, for an input dynamic range of 400 differential mVpp and a 1% distortion of the current output from the transconductor stage, the product gmbip*R should correspond to a value of six, meaning that the differential transconductor has a degeneration of 150 mV relative to the bipolar transistor. When minimal capacitances C1, C2, i.e. capacitances in the picoFarad range, are used in order to have the parasitic capacitances adequately controlled, currents in the mA range must be used for each differential transconductor to provide comparable time constants with those normally used in such applications. Lastly, it must be noted that the differential transconductor distortion is unrelated to the frequency of the signals being processed. That is, the current within the bipolar transistors incorporated into the differential transconductors is modulated, even for low frequency signals, to produce a variation in their transconductance values and, hence, a distortion in the output current. The underlying technical problem of this invention is to provide a biquadratic cell structure adapted to form continual time filters having such structural and functional features as to decrease the power dissipation through the filter, while retaining the performance level, in terms of dynamic speed, of the processed signal and the total harmonic distortion (THD) of the prior art filter. Summary of the Invention The solution idea on which the instant invention is based is one of providing a biquadratic cell which has a symmetrical structure and a smaller number of bias branches than conventional ones, and at the same time reducing the current flowed through each bias branch. The problem is solved by a biquadratic cell according to Claim 1 and following. The features and advantages of the invention will be apparent from the following description of embodiments thereof, given by way of illustration and not of limitation with reference to the accompanying drawings. Brief Description of the Drawings In the drawings: Figure 1 shows schematically a biquadratic elementary cell of the low-pass type, according to the prior art; Figure 2 shows schematically a differential transconductor included in the biquadratic cell of Figure 1; Figure 3 shows schematically a biquadratic cell according to this invention; Figure 4 shows a modified embodiment of the biquadratic cell of Figure 3; and Figures 5A and 5B are respective frequency response plots for a capacitively and resistively damped integrator. Detailed Description Referring in particular to Figure 3, generally and schematically shown at 30 is a biquadratic cell according to the invention. This cell is particularly, though not exclusively, intended for implementing programmable time-continous analog filters. The cell 30 is placed between a first supply voltage reference Vdd and a second voltage reference, specifically a signal ground GND, and is a symmetrical structure about an imaginary axis A-A. The cell 30 can be viewed as being split, at this axis A-A, into two half-cells 31 and 31', which are identical with each other and respectively connected to a pair of input terminals I31, I31', and to a first pair O31, O31' and a second pair O32, O32' of output terminals of the biquadratic cell 30. The first half-cell 31 comprises a first bipolar transistor T31, placed between the supply voltage reference Vdd and a first circuit node X31 and having a base terminal connected to the input terminal I31. The first circuit node X31 is also connected to the ground GND through a first bias current generator G31, and to a second circuit node X32 through a first resistor R31, the second circuit node X32 being further connected to the ground GND through a second bias current generator G32. The first half-cell 31 comprises a second bipolar transistor T32, which is placed between the second circuit node X32 and the second output terminal O32 and has a base terminal connected to the first output terminal O31, and comprises a third bipolar transistor T33, which is placed between the first output terminal O31 and a third circuit node X33 being also connected to the ground GND through a third bias current generator G33. The first O31 and second O32 output terminals are further connected to the supply voltage reference Vdd through fourth G34 and fifth G35 bias current generators. The first half-cell 31 also comprises first C31 and second C32 integration capacitors, respectively connected between the first O31 and second O32 output terminals and the ground GND. The structure of the second half-cell 31' is quite identical and symmetrical with that of the first half-cell 31 just described; accordingly, not to overburden this description, no detailed description thereof will be provided herein, except that corresponding elements to those of the first half-cell 31 are denoted in the figure by the same references numerals and characters with a prime (') added. The first half-cell 31 and second half-cell 31' are suitably connected to each other. In particular, the collector terminal of the second bipolar transistor T32 in the first half-cell 31, corresponding to the second output terminal O32 of the cell 30, is connected to the base terminal of the third bipolar transistor T33' of the second half-cell 31', and conversely, the collector terminal of the second bipolar transistor T32' in the second half-cell 31', corresponding to the second output terminal O32' of the cell 30, is connected to the base terminal of the third bipolar transistor T33 in the first half-cell 31. In addition, the emitter terminal of the third bipolar transistor T33 in the first half-cell 31, corresponding to the third circuit node X33, is connected to the emitter terminal of the second bipolar transistor T32'in the second half-cell 31', corresponding to the second circuit node X32', through a second resistor R32, and conversely, the emitter terminal of the third bipolar transistor T33' in the second half-cell 31', corresponding to the third circuit node X33', is connected to the emitter terminal of the second bipolar transistor T32 in the first half-cell 31, corresponding to the second circuit node X32, through a second resistor R32'. The main features of the resulting biquadratic cell 30 can be summarized as follows: (i) each half-cell, 31 and 31', has three bias branches only, compared to the four required to implement a corresponding half-cell according to the prior art; (ii) the bias currents flowed through the bipolar components of the biquadratic cell 30 can be reduced substantially compared to the prior art; and (iii) the transistor bias currents are only modulated by the high-frequency components (therefore, already filtered by the cell itself) of the input signal. Feature (ii) is provided by the adding together of the currents for the various transconductors being effected at the emitter terminals, rather than the collector terminals as in a biquadratic cell according to the prior art, so that the differential transconductors of the biquadratic cell will only be charged capacitively. The transfer function of the biquadratic cell 30 in Figure 3, as simplified by putting gmbip*R>>1, is given by: vy vin = 1 s 2 · C 1 · C 2 · R 1 · R 2 + s · R 1 · ( C 2 - C 1 ) + 1 vx vin = 1 + s · R 2 · C 1 s 2 · C 1 · C 2 · R 1 · R 2 + s · R 1 · ( C 2 - C 1 ) + 1 where, vin is the input voltage received at the input terminals I31, I31'; vx,vy are the output voltages respectively delivered on the first O31, O31' and second O32, O32' output terminals; C 1 ,C 2 are the respective values of the first C31, C31' and second C32, C32' capacitors; and R 1 ,R 2 are the values of the first R31, R31' and second R32, R32' resistors, respectively. Where gmbip*R is not much greater than one, the transfer function becomes complicated, but remains a transfer function of a biquadratic cell with outputs of the low-pass type, like that defined by the above relationships (2.1) and (2.2). However, the biquadratic cell 30 has a problem from the presence of a positive common mode feedback, which restricts its use to applications having a high common mode gain and a wide band. Shown in Figure 4 is a biquadratic cell 40 which overcomes the problems of the biquadratic cell 30, but still enjoys its beneficial features by merely providing for a partial modification of its circuit topology. For this biquadratic cell 40, all the corresponding elements to those of the cell 30 are denoted in Figure 4 by the same reference numerals and characters as in Figure 3. Compared to the cell 30, the first resistors R31 and R31' are respectively placed between the emitter terminals of the first T31, T31' and third T33, T33' bipolar transistors, while the second resistors R32 and R32' are placed, in series with each other, between the emitter terminals of the second bipolar transistors T32, T32'. In addition, the biquadratic cell 40 comprises respective damping capacitors CD, CD' connected between the emitter terminals of the third bipolar transistors T33, T33' and the ground GND. In particular, the biguadratic cell 40 will effect, in operation, a capacitive rather than resistive damping, through the capacitors CD and CD' having the same value Cd. These capacitors actually add a zero to the frequency gm/(2π*Cd) in the frequency response of one of the transconductors (having a transconductance gm). The resulting integrator shows a phase delay at its unit gain frequency, equal to gm/(2π*Cd). This phase delay corresponds to the phase delay of a transconductor having a transconductance gm and being charged from a resistor with a value of 1/gm3, of the type shown in Figure 1 in relation to the state of the art, so long as the following relationship holds: gm gm3 = C Cd Shown in Figure 5A is a plot of the frequency response for a capacitively damped integrator according to the invention, and Figure 5B shows a plot of the frequency response for a resistively damped integrator according to the prior art. The transfer function of the biquadratic cell 40 in Figure 4, as simplified by putting gmbip*R>>1, is given by: vy vin = -1 s 2 · C 1 · C 2 · R 1 · R 2 + s · R 1 · C D + 1 vx vin = s · R 2 · C 1 s 2 · C 1 · C 2 · R 1 · R 2 + s · R 1 · C D + 1 where, C D is the value of the damping capacitors CD and CD'. Thus, the biquadratic cell 40 has one output of the low-pass type (O31, O31') and another of a band-pass type (O32, O32'). A simulation carried out by the Applicant, using a biquadratic cell such as cell 40 in Figure 4 to implement a low-pass filter of the seventh order, with a frequency band of 50 MHz, revealed a reduction in consumption three times as great as that of conventional filters, while retaining the same harmonic distortion performance for a given dynamic range of the input signal and a given bandwidth. Modifications and changes may be made unto the biquadratic cell of this invention in an obvious manner for the skilled ones, without departing from the protection scope of the instant Application. For example, all of the points raised in connection with filters for reading data from bulk memories would apply in practice to any integrated electronic filters of the gm/C type.

Claims: An elementary biquadratic cell (30, 40) for programmable time-continous analog filters, said cell being placed between a first supply voltage reference (Vdd) and a second voltage reference (GND) and having at least one pair of input terminals (I31,I31') and first (O31,O31') and second (O32,O32') pairs of output terminals, said cell being formed by a pair of half-cells (31,31') which are are structurally identical with each other and comprise at least a first transistor (T31,T31') placed between the first (Vdd) and the second (GND) voltage reference and a base terminal connected to a respective one of the input terminals (I31,I31'), characterized in that each half-cell (31,31') further comprises second (T32,T32') and third (T33,T33') transistors placed between the first (Vdd) and second (GND) voltage references, the second transistor (T32,T32') having a base terminal connected to the first output terminal (O31,O31') of the first pair of output terminals and a collector terminal connected to the second output terminal (O32,O32') of the second pair of output terminals; the third transistor (T33,T33') having a collector terminal connected to the first output terminal (O31,O31') of the first pair of output terminals ad a base terminal connected to the second output terminal (O32',O32) of the second pair of output terminals of the other half-cell;said first transistor (T31,T31') having its emitter terminal (X31,X31') connected to the emitter terminal (X32,K32') of the second transistor (T32,T32') through a first resistor (R31,R31');said second transistor (T32,T32') having its emitter terminal (X32,X32') connected to the emitter terminal (X33,X33') of the third transistor (T33',T33) of the other half-cell through a second resistor (R32,R32'). An elementary biquadratic cell (40) according to Claim 1, characterized in that the third transistor (T33,T33') has its emitter terminal (X31,X31') connected to the second voltage reference (GND) through a damping capacitor (CD,CD'). An elementary biquadratic cell (30) according to Claim 1, characterized in that the first (O31,O31') and second (O32,O32') output terminals are connected to the second voltage reference (GND), through respective integration capacitors (C32,C32' and C31,C31'), and to the first voltage reference (Vdd) through respective bias current generators (G34,G34' and G35,G35'). An elementary biquadratic cell (30) according to Claim 1, characterized in that the first (T31,T31'), second (T32,T32') and third (T33,T33') transistors have their emitter terminals further connected to the second voltage reference (GND) through respective bias current generators (G31,G31'-G32, G32'-G33,G33'). An elementary biquadratic cell (40) according to Claim 2, characterized in that the first (O31,O31') and second (O32,O32') output terminals are connected to the second voltage reference (GND), through respective integration capacitors (C32,C32' and C31,C31'), and to the first voltage reference (Vdd) through respective bias current generators (G34,G34' and G35,G35'). An elementary biquadratic cell (40) according to Claim 2, characterized in that the first (T31,T31'), second (T32,T32') and third (T33,T33') transistors have their emitter terminals further connected to the second voltage reference (GND) through respective bias current generators (G31,G31'-G32, G32'-G33,G33').

Inventors: BIETTI IVAN, CASTELLO RINALDO, CAZZANIGA MARCO, REZZI FRANCESCO, BIETTI,IVAN, CASTELLO, RINALDO, CAZZANIGA, MARCO, REZZI, FRANCESCO
Applicants: ST MICROELECTRONICS SRL, STMICROELECTRONICS S.R.L.
Application year: 1996
Main classification: H03H11
IPC codes: H03H11