Speech decoding method and portable terminal apparatus

Abstract: A speech decoding method and apparatus for decoding encoded speech signals and subsequently post-filtering the decoded signals. The filter coefficient of a spectral shaping filter 440 in a post-filter fed with an encoded and subsequently decoded speech signal is updated with a sub-frame period, while the gain of a gain adjustment circuit 443 for correcting gain changes caused by the spectral shaping is updated with a frame period which is eight times as long as the sub-frame period. This achieves switching of the filter coefficient which is changed smoothly with a higher follow-up speed, while suppressing level changes otherwise caused by frequent gain switching. The result is improved characteristics of a post filter used for spectral shaping of a decoded signal supplied from the signal decoder and more effective post-filter processing.

Description: This invention relates to a speech decoding method and apparatus for decoding and subsequently post-filtering input speech signals. There have hitherto been known a variety of encoding methods for encoding an audio signal (inclusive of speech and acoustic signals) for compression by exploiting statistic properties of the signals in the time domain and in the frequency. domain and psychoacoustic characteristics of the human ear. The encoding method may roughly be classified into time-domain encoding, frequency domain encoding and analysis/synthesis encoding. Examples of the high-efficiency encoding of speech signals include sinusoidal analysis encoding, such as harmonic encoding, multi-band excitation (MBE) encoding, sub-band coding (SBC), linear predictive coding (LPC), discrete cosine transform (DCT), modified DCT (MDCT) and fast Fourier transform (FFT). Post-filters are sometimes used after decoding these encoded signals for spectral shaping and improving the psychoacoustic signal quality. Thus, such known post filters comprise: spectral shaping filtering means for spectrally shaping a decoded input signal with a filter coefficient updated periodically; and gain adjustment means fed with an output of the spectral shaping filtering means and arranged to perform gain adjustment for correcting gain changes caused by the spectral shaping filter means using a gain which is updated periodically, wherein the gain and filter coefficients are updated with the same period. If, in updating the filter characteristics responsive to an input, the updating period is prolonged, the post filter characteristics cannot follow the short-term changes in the speech spectrum, so a smooth and optimum improvement in signal quality cannot be achieved. Moreover, if the updating period is short, level changes become severe such that a click noise tends to be produced. US-A-5,339,384, on which the two part form of claim 1 is based, discloses a speech decoding apparatus in which an encoded speech signal is input decoded and subsequently post-filtered, the apparatus including: spectral shaping filtering means for spectrally shaping a decoded input signal with a filter coefficient updated periodically; and gain adjustment means fed with an output of said spectral shaping filtering means arranged to perform gain adjustment for correcting gain changes caused by said spectral shaping filter means using a gain which is updated periodically, wherein the filter coefficients are updated with a period of one frame of 20 samples (2.5ms) and the gain is updated with a period of one sample (0.125ms) by low-pass filter of a gain calculation updated with a period of one vector of five samples (0.625ms). It is therefore an object of the present invention to provide a speech decoding method whereby a satisfactory decoded output and a high quality playback sound can be produced even if the method is used for decoding the speech encoded with a smaller number of bits. According to the first aspect of the present invention, there is provided a speech decoding method in which an encoded speech signal is input decoded and subsequently post-filtered, the method including: performing spectral shaping filtering of the decoded input signal using filter coefficients which are updated periodically; and performing gain adjustment using a gain which is updated periodically for correcting gain changes caused by said spectral shaping filtering characterised in that the gain is updated with a second period which is longer than the first period with which the filter coefficients are updated. According to the second aspect of the present invention there is provided a speech decoding apparatus in which an encoded speech signal is input decoded and subsequently post-filtered, the apparatus including: spectral shaping filtering means for spectrally shaping a decoded input signal with a filter coefficient updated periodically; and gain adjustment means fed with an output of said spectral shaping filtering means arranged to perform gain adjustment for correcting gain changes caused by said spectral shaping filter means using a gain which is updated periodically; characterised in that the gain adjustment means is arranged to updated the gain with a second period which is longer than the first period with which the filter coefficients are updated. In this case, by relatively shortening the first period as the updating period of the filter coefficient of the spectral shaping filter, and by relatively elongating the second period as the gain updating the period for gain adjustment, it becomes possible to effect switching of the smoothly changed filter coefficient with a high follow-up rate and to suppress sudden level changes otherwise caused by frequent gain switching. Specifically, if the longer updating period of the filter coefficient of the spectral shaping filter in a post filter is used, post-filter characteristics cannot cope with short-term changes in the speech spectrum, thereby deteriorating the quality of the output speech. In this consideration, it may be contemplated to update the filter coefficient with a short period. However, if the gain value for adjustment is updated in a correspondingly short period, there may be occasion wherein the gain value for gain adjustment is significantly changed within the one-pitch period given the pitch and peak phase conditions, thus producing click noise. In this consideration, the filter coefficient updating period and the gain value updating period are relatively shortened and elongated, respectively, for suppressing gain variations for realizing optimum post filtering. The present invention will be more clearly understood from the following description, given by way of example only, with reference to the accompanying drawings in which: Fig. 1 is a block diagram showing a basic structure of an embodiment of a speech encoding apparatus for producing the encoded speech entering a speech decoding apparatus according to the present invention. Fig.2 is a block diagram showing a basic structure of an embodiment of a speech decoding apparatus for carrying out the speech decoding method according to the present invention. Fig.3 is a block diagram showing a detailed structure of the speech signal encoding apparatus shown in Fig. 1. Fig.4 is a block diagram showing a detailed structure of the speech signal decoding apparatus according to the present invention. Fig.5 shows ten-order linear spectral pair (LSP) derived from α-parameter obtained by 10-order LPC analysis. Fig.6 illustrates the manner of gain change from an unvoiced (UV) frame to a voiced (V) frame. Fig.7 illustrates interpolation of the spectrum or the waveform synthesized on the frame basis. Fig.8 illustrates overlap at a junction between a voiced (V) frame and an unvoiced (UV) frame. Fig.9 illustrates noise addition at the time of synthesis of the voiced sound. Fig.10 illustrates an example of amplitude calculations of the noise summed at the time of synthesis of the voiced sound. Fig.11 illustrates a typical structure of a post-filter. Fig.12 illustrates the post-filter coefficient updating period and the gain updating period. Fig.13 illustrates a connection operation at a frame boundary portion of the filter coefficient and the post filter gain. Fig.14 is a block diagram showing the structure of the transmitting side of a portable terminal employing the speech signal encoding apparatus of the present invention. Fig.15 is a block diagram showing the structure of the receiving side of the portable terminal employing the speech signal decoding apparatus according to the present invention. Prior to description of the preferred embodiment of the invention, the speech encoding apparatus and the speech decoding apparatus, as an example of the speech codec according to the present invention, will be explained by referring to the drawings. Fig.1 shows a basic structure of a speech encoding apparatus (encoder). The basic concept of the speech signal encoder of Fig. 1 is that the encoder has a first encoding unit 110 for finding short-term prediction residuals, such as linear prediction encoding (LPC) residuals, of the input speech signal for performing sinusoidal analysis encoding, such as harmonic coding, and a second encoding unit 120 for encoding the input speech signals by waveform coding exhibiting phase reproducibility, and that the first encoding units 110, 120 are used for encoding the voiced portion and unvoiced portion of the input signal, respectively. The first encoding unit 110 has a constitution of encoding the LPC residuals with sinusoidal analytic encoding such as harmonics encoding or multi-band encoding (MBE).The second encoding unit 120 has a constitution of code excitation linear prediction (CELP) employing vector quantization by a closed loop search for an optimum vector employing an analysis by synthesis method. In the embodiment, the speech signal supplied to the input terminal 101 is sent the inverse LPC filter 111 and an LPC analysis/quantization unit 113 of the first encoding unit 110. The LPC coefficient obtained from the LPC analysis/quantization unit 113 or the so-called α-parameter is sent to the inverse LPC filter 111 for taking out the linear prediction residuals (LPC residuals) of the input speech signals by the inverse LPC filter 111. From the LPC analysis/quantization unit 113, a quantization output of the linear spectral pairs (LSP) is taken out as later explained and sent to an output terminal 102. The LPC residuals from the inverse LPC filter 111 are sent to a sinusoidal analysis encoding unit 114. The sinusoidal analysis encoding unit 114 performs pitch detection, spectral envelope amplitude calculations and V/UV discrimination by a voiced (V)/ unvoiced (UV) discrimination unit 115. The spectral envelope amplitude data from the sinusoidal analysis encoding unit 114 are sent to the vector quantization unit 116. The codebook index from the vector quantization unit 116, as a vector quantization output of the spectral envelope, is sent via a switch 117 to an output terminal 103, while an output of the sinusoidal analysis encoding unit 114 is sent via a switch 118 to an output terminal 104. The V/UV discrimination output from the V/UV discrimination unit 115 is sent to an output terminal 105 and to the switches 117, 118 as switching control signals. For the voiced (V) signal, the index and the pitch are selected so as to be taken out at the output terminals 103, 104. In the present embodiment, the second encoding unit 120 of Fig.1 has a code excitation linear prediction (CELP) encoding configuration, and performs vector quantization of the time-domain waveform employing the closed-loop search by the analysis by synthesis method in which an output of a noise codebook 121 is synthesized by a weighted synthesis filter 122, the resulting weighted speech is sent to a subtractor 123, where an error between the weighted speech and the speech signal supplied to the input terminal 101 and thence passed through a perceptually weighted filter 125 is taken out and sent to a distance calculation circuit 124 in order to perform distance calculations, while a vector which minimizes the error is searched by the noise codebook 121. This CELP encoding is used for encoding the unvoiced portion as described above. The codebook index as the UV data from the noise codebook 121 is taken out at an output terminal 107 via a switch 127 which is turned on when the results of V/UV discrimination from the V/UV discrimination unit 115 indicates an unvoiced (UV) sound. Fig.2 is a block diagram showing the basic structure of a speech signal decoder, as a counterpart device of the speech signal encoder of Fig. 1, for carrying out the speech decoding method according to the present invention. Referring to Fig.2, a codebook index as a quantization output of the linear spectral pairs (LSPs) from the output terminal 102 of Fig.1 is supplied to an input terminal 202. Outputs of the output terminals 103, 104 and 105 of Fig.1, that is the index data, pitch and the V/UV discrimination output as the envelope quantization outputs, are supplied to input terminals 203 to 205, respectively. The index data as data for the unvoiced data are supplied from the output terminal 107 of Fig.1 to an input terminal 207. The index as the quantization output of the input terminal 203 is sent to an inverse vector quantization unit 212 for inverse vector quantization to find a spectral envelope of the LPC residues which is sent to a voiced speech synthesizer 211. The voiced speech synthesizer 211 synthesizes the linear prediction encoding (LPC) residuals of the voiced speech portion by sinusoidal synthesis. The voiced speech synthesizer 211 is also fed with the pitch and the V/UV discrimination output from the input terminals 204, 205. The LPC residuals of the voiced speech from the voiced speech synthesis unit 211 are sent to an LPC synthesis filter 214. The index data of the UV data from the input terminal 207 is sent to an unvoiced sound synthesis unit 220 where reference is had to the noise codebook for taking out the LPC residuals of the unvoiced portion. These LPC residuals are also sent to the LPC synthesis filter 214. In the LPC synthesis filter 214, the LPC residuals of the voiced portion and the LPC residuals of the unvoiced portion are processed by LPC synthesis. Alternatively, the LPC residuals of the voided portion and the LPC residuals of the unvoiced portion summed together may be processed with LPC synthesis. The LSP index data from the input terminal 202 is sent to the LPC parameter reproducing unit 213 where α-parameters of the LPC are taken out and sent to the LPC synthesis filter 214. The speech signals synthesized by the LPC synthesis filter 214 are taken out at an output terminal 201. Referring to Fig.3, a more detailed structure of a speech signal encoder shown in Fig.1 is now explained. In Fig.3, the parts or components similar to those shown in Fig.1 are denoted by the same reference numerals. In the speech signal encoder shown in Fig.3, the speech signals supplied to the input terminal 101 are filtered by a high-pass filter 109 for removing signals of an unneeded range and thence supplied to an LPC analysis circuit 132 of the LPC analysis/quantization unit 113 and to the inverse LPC filter 111. The LPC analysis circuit 132 of the LPC analysis/quantization unit 113 applies a Hamming window, with a length of the input signal waveform on the order of 256 samples as a block, and finds a linear prediction coefficient, that is a so-called α-parameter, by the self-correlation method. The framing interval as a data outputting unit is set to approximately 160 samples. If the sampling frequency fs is 8 kHz, for example, a one-frame interval is 20 msec for 160 samples. The α-parameter from the LPC analysis circuit 132 is sent to an α-LSP conversion circuit 133 for conversion into line spectra pair (LSP) parameters. This converts the α-parameter, as found by direct type filter coefficient, into for example, ten, that is five pairs of the LSP parameters. This conversion is carried out by, for example, the Newton-Rhapson method. The reason the α-parameters are converted into the LSP parameters is that the LSP parameter is superior in interpolation characteristics to the α-parameters. The LSP parameters from the α-LSP conversion circuit 133 are matrix- or vector quantized by the LSP quantizer 134. It is possible to take a frame-to-frame difference prior to vector quantization, or to collect plural frames in order to perform matrix quantization. In the present case, two frames (20 msec) of the LSP parameters, calculated every 20 msec, are collected and processed with matrix quantization and vector quantization. The quantized output of the quantizer 134, that is the index data of the LSP quantization, are taken out at a terminal 102, while the quantized LSP vector is sent to an LSP interpolation circuit 136. The LSP interpolation circuit 136 interpolates the LSP vectors, quantized every 20 msec or 40 msec, in order to provide an eight-fold rate. That is, the LSP vector is updated every 2.5 msec. The reason is that, if the residual waveform is processed with the analysis/synthesis by the harmonic encoding/decoding method, the envelope of the synthetic waveform presents an extremely sooth waveform, so that, if the LPC coefficients are changed abruptly every 20 msec, a foreign noise is likely to be produced. That is, if the LPC coefficient is changed gradually every 2.5 msec, such foreign noise may be prevented from occurrence. For inverse filtering of the input speech using the interpolated LSP vectors produced every 2.5 msec, the LSP parameters are converted by an LSP to α conversion circuit 137 into α-parameters as coefficients of e.g., ten-order direct type filter. An output of the LSP to α conversion circuit 137 is sent to the LPC inverse filter circuit 111 which then performs inverse filtering for producing a smooth output using an α-parameter updated every 2.5 msec. An output of the inverse LPC filter 111 is sent to an orthogonal transform circuit 145, such as a DCT circuit, of the sinusoidal analysis encoding unit 114, such as a harmonic encoding circuit. The α-parameter from the LPC analysis circuit 132 of the LPC analysis/quantization unit 113 is sent to a perceptual weighting filter calculating circuit 139 where data for perceptual weighting is found. These weighting data are sent to a perceptual weighting vector quantizer 116, perceptual weighting filter 125 of the second encoding unit 120 and the perceptual weighted synthesis filter 122. The sinusoidal analysis encoding unit 114 of the harmonic encoding circuit analyzes the output of the inverse LPC filter 111 by a method of harmonic encoding. That is, pitch detection, calculations of the amplitudes Am of the respective harmonics and voiced (V)/ unvoiced (UV) discrimination are carried out and the numbers of the amplitudes Am or the envelopes of the respective harmonics, varied with the pitch, are made constant by dimensional conversion. In an illustrative example of the sinusoidal analysis encoding unit 114 shown in Fig.3, commonplace harmonic encoding is used. In particular, in multi-band excitation (MBE) encoding, it is assumed in modeling that voiced portions and unvoiced portions are present in the frequency area or band at the same time point (in the same block or frame). In other harmonic encoding techniques, it is uniquely judged whether the speech in one block or in one frame is voiced or unvoiced. In the following description, a given frame is judged to be UV if the totality of the band is UV, insofar as the MBE encoding is concerned. The open-loop pitch search unit 141 and the zero-crossing counter 142 of the sinusoidal analysis encoding unit 114 of Fig.3 is fed with the input speech signal from the input terminal 101 and with the signal from the high-pass filter (HPF) 109, respectively. The orthogonal transform circuit 145 of the sinusoidal analysis encoding unit 114 is supplied with LPC residuals or linear prediction residuals from the inverse LPC filter 111. The open loop pitch search unit 141 takes the LPC residuals of the input signals to perform relatively rough pitch search by open loop. The extracted rough pitch data is sent to a fine pitch search unit 146 by closed loop as later explained. From the open loop pitch search unit 141, the maximum value of the normalized self correlation r(p), obtained by normalizing the maximum value of the self-correlation of the LPC residuals along with the rough pitch data, are taken out along with the rough pitch data so as to be sent to the V/UV discrimination unit 115. The orthogonal transform circuit 145 performs orthogonal transform, such as discrete Fourier transform (DFT) for converting the LPC residuals on the time axis into spectral amplitude data on the frequency axis. An output of the orthogonal transform circuit 145 is sent to the fine pitch search unit 146 and a spectral evaluation unit 148 for evaluating the spectral amplitude or envelope. The fine pitch search unit 146 is fed with relatively rough pitch data extracted by the open loop pitch search unit 141 and with frequency-domain data obtained by DFT by the orthogonal transform unit 145. The fine pitch search unit 146 swings the pitch data by ± several samples, at a rate of 0.2 to 0.5, centered about the rough pitch value data, in order to arrive ultimately at the value of the fine pitch data having an optimum decimal point (floating point). The analysis by synthesis method is used as the fine search technique for selecting a pitch so that the power spectrum will be closest to the power spectrum of the original sound. Pitch data from the closed-loop fine pitch search unit 146 is sent to an output terminal 104 via a switch 118. In the spectral evaluation unit 148, the amplitude of each harmonics and the spectral envelope as the sum of the harmonics are evaluated based on the spectral amplitude and the pitch as the orthogonal transform output of the LPC residuals and sent to the fine pitch search unit 146, V/UV discrimination unit 115 and the perceptually weighted vector quantization unit 116. The V/UV discrimination unit 115 discriminates V/UV of a frame based on an output of the orthogonal transform circuit 145, an optimum pitch from the fine pitch search unit 146, spectral amplitude data from the spectral evaluation unit 148, maximum value of the normalized self-correlation r(p) from the open loop pitch search unit 141 and the zero-crossing count value from the zero-crossing counter 142. In addition, the boundary position of the band-based V/UV discrimination for the MBE may also be used as a condition for V/UV discrimination. A discrimination output of the V/UV discrimination unit 115 is taken out at an output terminal 105. An output unit of the spectrum evaluation unit 148 or an input unit of the vector quantization unit 116 is provided with a data number conversion unit (a unit performing a sort of sampling rate conversion). The data number conversion unit is used for setting the amplitude data |Am| of an envelope taking into account the fact that the number of bands split on the frequency axis and the number of data differ with the pitch. That is, if the effective band is up to 3400 kHz, the effective band can be split into 8 to 63 bands depending on the pitch. The number of mMX + 1 of the amplitude data |Am|, obtained from band to band, is changed in a range from 8 to 63. Thus the data number conversion unit 119 converts the amplitude data of the variable number mMx + 1 to a pre-set number M of data, such as 44 data. The amplitude data or envelope data of the pre-set number M, such as 44, from the data number conversion unit, provided at an output unit of the spectral evaluation unit 148 or at an input unit of the vector quantization unit 116, are collected in terms of a pre-set number of data, such as 44 data, as units, by the vector quantization unit 116, by way of performing weighted vector quantization. This weight is supplied by an output of the perceptual weighting filter calculation circuit 139. The index of the envelope from the vector quantizer 116 is taken out by a switch 117 at an output terminal 103. Prior to weighted vector quantization, it is advisable to take inter-frame difference using a suitable leakage coefficient for a vector made up of a pre-set number of data. An illustrative arrangement for data number conversion for providing a constant number of data of the amplitude of the spectral envelope on an output side of the spectral evaluating unit 148 or on an input side of the vector quantization unit 116 is explained. A variety of methods may be conceived for such data number conversion. In the present embodiment, dummy data interpolating the values from the last data in a block to the first data in the block or other pre-set data such as data repeating the last data or the first data in a block are appended to the amplitude data of one block of an effective band on the frequency axis for enhancing the number of data to N F , amplitude data equal in number to Os times, such as eight times, are found by Os-fold, such as eight-fold oversampling of the limited bandwidth type by, for example, an FIR filter. The ((mMx + 1) × Os amplitude data are linearly interpolated for expansion to a larger N M number, such as 2048. This N M data is sub-sampled for conversion to the above-mentioned pres-set number M of data, such as 44 data. The second encoding unit 120 is explained. The second encoding unit 120 has a so-called CELP encoding structure and is used in particular for encoding the unvoiced portion of the input speech signal. In the CELP encoding structure for the unvoiced portion of the input speech signal, a noise output, corresponding to the LPC residuals of the unvoiced sound as a representative value output of the noise codebook, or a so-called stochastic codebook 121, is sent via a gain control circuit 126 to a perceptually weighted synthesis filter 122. The weighted synthesis filter 122 LPC synthesizes the input noise and sends the produced weighted unvoiced signal to the subtractor 123. The subtractor 123 is fed with a signal supplied from the input terminal 101 via an high-pass filter (HPF) 109 and perceptually weighted by a perceptual weighting filter 125. The difference or error between the signal and the signal from the synthesis filter 122 is taken out. Meanwhile, a zero input response of the perceptually weighted synthesis filter is previously subtracted from an output of the perceptual weighting filter output 125. This error is fed to a distance calculation circuit 124 for calculating the distance. A representative vector value which will minimize the error is searched in the noise codebook 121. The above is the summary of the vector quantization of the time-domain waveform employing the closed-loop search in turn employing the analysis by synthesis method. As data for the unvoiced (UV) portion from the second encoder 120 employing the CELP coding structure, the shape index of the codebook from the noise codebook 121 and the gain index of the codebook from the gain circuit 126 are taken out. The shape index, which is the UV data from the noise codebook 121, and the gain index, which is the UV data of the gain circuit 126, are sent via a switch 127 g to an output terminal 107 g. These switches 127s, 127 g and the switches 117, 118 are turned on and off depending on the results of V/UV decision from the V/UV discrimination unit 115. Specifically, the switches 117, 118 are turned on, if the results of V/UV discrimination of the speech signal of the frame currently transmitted indicates voiced (V), while the switches 127s, 127 g are turned on if the speech signal of the frame currently transmitted is unvoiced (UV). Fig.4 shows a more detailed structure of a speech signal decoder shown in Fig.2. In Fig.4, the same numerals are used to denote the opponents shown in Fig.2. In Fig.4, a vector quantization output of the LSP corresponding to the output terminal 102 of Figs.1 and 3, that is the codebook index, is supplied to an input terminal 202. The LSP index is sent to the inverse vector quantizer 231 of the LSP for the LPC parameter reproducing unit 213 so as to be inverse vector quantized to line spectral pair (LSP) data which are then supplied to LSP interpolation circuits 232, 233 for interpolation. The resulting interpolated data is converted by the LSP to α conversion circuits 234, 235 to α parameters which are sent to the LPC synthesis filter 214. The LSP interpolation circuit 232 and the LSP to α conversion circuit 234 are designed for voiced (V) sound, while the LSP interpolation circuit 233 and the LSP to α conversion circuit 235 are designed for unvoiced (UV) sound. The LPC synthesis filter 214 separates the LPC synthesis filter 236 of the voiced speech portion from the LPC synthesis filter 237 of the unvoiced speech portion. That is, LPC coefficient interpolation is carried out independently for the voiced speech portion and the unvoiced speech portion for prohibiting ill effects which might otherwise be produced in the transition portion from the voiced speech portion to the unvoiced speech portion or vice versa by interpolation of the LSPs of totally different properties. To an input terminal 203 of Fig.4 is supplied code index data corresponding to the weighted vector quantized spectra envelope Am corresponding to the output of the terminal 103 of the encoder of Figs.1 and 3. To an input terminal 204 is supplied pitch data from the terminal 104 of Figs.1 and 3 and, to an input terminal 205 is supplied V/UV discrimination data from the terminal 105 of Figs.1 and 3. The vector-quantized index data of the spectral envelope Am from the input terminal 203 is sent to an inverse vector quantizer 212 for inverse vector quantization where an inverse conversion with respect to the data number conversion is carried out. The resulting spectral envelope data is sent to a sinusoidal synthesis circuit 215. If the inter-frame difference is found prior to vector quantization of the spectrum during encoding, inter-frame difference is decoded after inverse vector quantization for producing the spectral envelope data. The sinusoidal synthesis circuit 215 is fed with the pitch from the input terminal 204 and the V/UV discrimination data from the input terminal 205. From the sinusoidal synthesis circuit 215, LPC residual data corresponding to the output of the LPC inverse filter 111 shown in Figs.1 and 3 are taken out and sent to an adder 218. The envelop data of the inverse vector quantizer 212 and the pitch and the V/UV discrimination data from the input terminals 204, 205 are sent to a noise synthesis circuit 216 for noise addition for the voiced portion (V). An output of the noise synthesis circuit 216 is sent to an adder 218 via a weighted overlap-add circuit 217. That is, such noise is added to the voiced portion of the LPC residual signals which takes into account the fact that, if the excitation as an input to the LPC synthesis filter of the voiced sound is produced by sine wave synthesis, stuffed feeling is produced in the low-pitch sound such as male speech, and the sound quality is abruptly changed between the voiced sound and the unvoiced sound thus producing an unnatural hearing feeling. Such noise takes into account the parameters concerned with speech encoding data, such as pitch, amplitudes of the spectral envelope, maximum amplitude in a frame or the residual signal level, in connection with the LPC synthesis filter input of the voiced speech portion, that is excitation. An addition output of the adder 218 is sent to a synthesis filter 236 for the voiced sound of the LPC synthesis filter 214 where LPC synthesis is carried out to form time waveform data which then is filtered by a post-filter 238v for the voiced speech and sent to the adder 239. The post-filter 238v for voiced sound shortens the update period of the filter coefficient of the internal spectral shaping filter to 20 samples or 2.5 msec, while elongating the gain update period of the gain adjustment circuit to 160 samples or 20 msec, as will be explained subsequently. The shape index and the gain index, as UV data from the output terminals 107s and 107 g of Fig.3, are supplied to the input terminals 207s and 207g of Fig.4, and thence supplied to the unvoiced speech synthesis unit 220. The shape index from the terminal 207s is sent to the noise codebook 221 of the unvoiced speech synthesis unit 220, while the gain index from the terminal 207g is sent to the gain circuit 222. The representative value output read out from the noise codebook 221 is a noise signal component corresponding to the LPC residuals of the unvoiced speech. This becomes a pre-set gain amplitude in the gain circuit 222 and is sent to a windowing circuit 223 so as to be windowed for smoothing the junction to the voiced speech portion. An output of the windowing circuit 223 is sent to a synthesis filter 237 for the unvoiced (UV) speech of the LPC synthesis filter 214 as an output of the unvoiced speech synthesis unit 220. The data sent to the synthesis filter 237 is processed with LPC synthesis to become time waveform data for the unvoiced portion. The time waveform data of the unvoiced portion is filtered by a post-filter 238u for the unvoiced portion before being sent to an adder 239. The post-filter 238u for unvoiced sound also shortens the update period of the filter coefficient of the internal spectral shaping filter to 20 samples or 2.5 msec, while elongating the gain update period of the gain adjustment circuit to 160 samples or 20 msec, as later explained. Alternatively, the updating frequency of the spectra shaping filter coefficient may be matched to that of the LPC synthesis filter for UV of the synthesis filter 237 insofar as the unvoiced speech is concerned. In the adder 239, the time waveform signal from the post-filter for the voiced speech 238v and the time waveform data for the unvoiced speech portion from the post-filter 238u for the unvoiced speech are added to each other and the resulting sum data is taken out at the output terminal 201. The detailed structure and operation of the speech signal decoder of Fig.4 are now explained. The LPC synthesis filter 214 is divided into the synthesis filter for voiced sound (V) 236 and the synthesis filter for unvoiced sound (UV) 237, as explained previously. That is, if the synthesis filter is not split and LSP interpolation is continuously performed without making distinction between V and UV every 20 samples, that is every 2.5 msec, the LSPs of totally different properties are interpolated at the V to UV and UV to V transient portions, so that the LPC of UV is used for the residuals of V and the LPC of V is used for the residuals of UV so that a foreign sound is produced. For avoiding these ill effects, the LPC synthesis filter is separated into a filter for V and a filter for UV and a filter interpolation for UV and LPC coefficient interpolation is performed independently for V and UV. The method for coefficient interpolation of the LPC filters 236, 237 is now explained. The LSP interpolation is switched depending on the V/UV state, as shown in Table 1. Hv(z) Huv(z) previous frame current frame previous frame current frame V to V transmitted LSP transmitted LSP equal interval LSP equal interval LSP V to UV transmitted LSP equal interval LSP equal interval LSP transmitted LSP UV to V equal interval LSP transmitted LSP transmitted LSP equal interval LSP UV to UV equal interval LSP equal interval LSP transmitted LSP transmitted LSP In Table 1, the equal interval LSP in case of 10-order LPC analysis means LSP associated with α-parameters for flat filter characteristics and gain qual to unity, that is α0 = 1, α1 = α2 = ...α10 = 0, that is LSPi = (π/11) × i, where 0 ≤ i ≤ 10. For the 10-order LPC analysis, that is 10-order LSP, the LSPs are equally arrayed at different positions obtained on equally dividing the interval between 0 and π into 11 and correspond to completely flat spectrum. The full-range gain of the synthesis filter presents minimum-through characteristics. Fig.6 schematically shows the manner of gain changes. That is, Fig.6 shows how the gain of 1/Huv(z) and the gain for 1/Hv(z) are changed during transition from the unvoiced (UV) portion to the voiced (V) portion. If the frame interval is 160 samples or 20 msec, the coefficient for 1/Hv(z) is interpolated every 2.5 msec or every 20 samples, while the coefficient for 1/Huv(z) is 10 msec (80 samples) and 5 msec (40 samples) for the bit rates of 2 kbps and 6 kbps, respectively. For UV, waveform matching is done with the aid of the analysis by synthesis method by the second encoding unit 120 on the encoder side, so that interpolation can be done with the LSPs of the neighboring UV portion instead of with the equal interval LSPs. In the UV encoding in the second encoding unit 120, the zero input response is set to zero by clearing the internal state of the weighted synthesis filter 122 of 1/A(z) at the transition portion from V to UV. Outputs of these LPC synthesis filters 236, 237 are sent to independently provided post-filters 238v, 238u. By post-filtering independently for V and UV, the intensity and frequency response of the post-filter are set to different values for V and UV. The windowing for the junction portion between the V and UV portions of the LPC residual signals, that is the excitation as an LPC synthesis filter input, is explained. This is performed by the sinusoidal synthesis circuit 215 of the voiced sound synthesis unit 211 and a windowing circuit 223 of the unvoiced sound synthesis unit 220. For the voiced (V) portion, since the spectral components are interpolated using the spectral components of the neighboring frames, all waveforms across the n'th and the (n+1)th frames can be produced, as shown in Fig.7. However, for a portion lying across the V and UV frames, such as the (n+1)th frame and (n+2)th frame in Fig.7, only data of ±80 samples in the frame are encoded and decoded. It is noted that 160 samples make up one frame interval. To this end, windowing is done beyond the center point CN between the frames on the V side, while it is done as far as the center point CN on the UV side, with an overlap at the connecting portion, as shown in Fig.8. The reverse operation is done on the UV to V transient portion. The windowing on the V side may be done as indicated by a broken line in Fig.8. The noise synthesis and noise addition for the voiced (V) portion is explained. To this end, using the noise synthesis circuit 216, weighted overlap circuit 217 and the adder 218 of Fig.4, the noise taking into account the following parameters is added to the voiced portion of the LPC residual signals for the excitation which becomes the LPC filter input of the voiced portion. These parameters include the pitch lag Pch, spectral amplitudes of the voiced sound Am[i], maximum spectral amplitude in the frame Amax and the level Lev of the residual signals. The pitch lag Pch is the number of samples in the pitch period for a pre-set sampling frequency fs, such as 8 kHz, while i in the spectral amplitude As[i] is an integer such that 0 < i < I, where I is the number of harmonics in the band of fs/2 (I = Pch/2). The processing by the noise synthesis circuit 216 is performed in a similar manner to the synthesis of the unvoiced sound for MBE. Fig.9 shows an illustrative example of the noise synthesis circuit 216. That is, in Fig.9, there is outputted from a Gaussian noise generator 401 the Gaussian noise corresponding to the time-domain white noise signal waveform windowed to a pre-set length of, for example, 256 samples, by a suitable window function, such as a Hamming window. This output signal is transformed by short-term Fourier transform (STFT) by a STFT unit 402 to produce a noise power spectrum on the frequency axis. The power spectrum from the STFT unit 402 is sent to a multiplier 403 for amplitude processing where it is multiplied with an output of the noise amplitude control circuit 410. An output of the multiplier 403 is sent to an inverse STFT unit 404 so as to be inverse STFTed for conversion to the time-domain signal using the phase of the original white noise. An output of the ISTFT unit 404 is sent to the weighting overlap-add circuit 217 of Fig.4. Instead of using the arrangement of the white noise generator 401 and the STFT unit 402, it is also possible to generate random numbers and to use them as real part or imaginary part or as the amplitude or phase of the white noise spectrum for processing, thereby omitting the STFT unit 402. The noise amplitude control circuit 410 has the basic structure as shown in Fig.10 and controls the multiplication coefficients of the multiplier 403, based on the spectral amplitude Am(i) for the voiced sound supplied from the dequantizer 212 for the spectral envelope shown in Fig.4 via terminal 411 and on the pitch lag Pch supplied from the input terminal 204 of Fig.4 via terminal 412, for finding the synthesized noise amplitude Am_ noise[i]. That is, in Fig.10, an output of a calculation circuit 416 for an optimum noise mix value fed with the spectral amplitude Am[i] and the pitch lag Pch, is weighted by a noise weighting circuit 417, and the resulting output is sent to a multiplier 418 for being multiplied with the spectral amplitude Am[i] for producing a noise amplitude Am noise[i]. A first illustrative example of noise synthesis and addition, in which the noise amplitude Am_ noise[i] becomes two of the above four parameters, namely the pitch lag Pch and the function f1(Pch,Am[i]) of the spectral amplitude Am[i], is now explained. Among the illustrative examples of these functions f1(Pch,Am[i]), there are f1(Pch,Am[i]) = 0 (0 < i < Noise_ b × I) f1(Pch,Am[i]) = Am[i] × noise_ mix (Noise_ b × I ≤ i < 1). The maximum value of noise_ mix is noise_ mix_ max, which is the clipping point. As an example, K = 0.02, noise_ mix_ max = 0.3 and Noise_ b = 0.7, where Noise_ b is a constant for determining in which partial portion of the entire area to begin to add the noise. In the present example, the noise is added beginning from 70% portion of the entire area, that is for a range from 4000×7 = 2800 Hz to 4000 Hz for fs = 8 kHz. A second illustrative example of noise synthesis and addition, in which the noise amplitude Am_ noise[i] becomes three of the above four parameters, namely the pitch lag Pch, spectral amplitude Am[i] and the function f2(Pch,Am[i],Amax) of the maximum spectral amplitude Amax, is now explained. Among the illustrative examples of these functions f2(Pch,Am[i],Amax), there are f2(Pch,Am[i],Amax) = 0 (0 < i < Noise_ b × I) f2(Pch,Am[i],Amax) = Am[i] × noise_ mix (Noise_ b × I ≤ i < I) noise_ max = K × Pch/ 2.0 It is noted that the maximum value of noise_ mix is noise_ mix max and, by way of examples, K = 0.02, noise_ mix_ max = 0.3 and Noise_ b = 0.7. Further, if Am[i] × noise_ mix > A max × C × noise_ mix, f2(Pch,Am[i],Amax) = A max × C × noise_ mix. Since the noise level can be prevented from being increased excessively by this condition, K and noise_ mix_ max can be enlarged further such that the noise level can be increased if the high-range level is also higher. A third illustrative example of noise synthesis and addition in which the noise amplitude Am_ noise[i] may become the function f3 (Pch,Am[i], Amax, Lev) of all of the above four parameters, is now explained. An illustrative example of such function f3 (Pch,Am[i],Amax,Lev) is basically the same as the function f2(Pch,SAm[i],Amax) of the above second illustrative example. However, the residual signal level Lev is the root mean square (rms) of the spectral amplitude Am[i], or the signal level as measured on the time axis. The difference of the present example from the second illustrative example lies in setting the values of K and noise mix_ max as the functions of Lev. That is, if Lev becomes smaller, the values of L and noise_ mix_ max may be set to higher values, whereas, if Lev is larger, the values of L and noise_ mix_ max may be set to lower values. Alternatively, the value of Lev may be set so as to be continuously inversely proportionate to these values. The post-filters 238v, 238u will be explained. Referring to Fig.11, showing a post-filter employed as the post filter 238v or 238u of Fig.4, a spectral shaping filter 440, used as an essential portion of the post-filter, is made up of a formant stressing filter 441 and a high-range stressing filter 442. An output of the spectral shaping filter 440 is sent to a gain adjustment circuit 443 for correcting gain changes caused by spectral shaping. A gain G of the gain adjustment circuit 443 is set by a gain control circuit 445 which compares an input x and an output y of the spectral shaping filter 440 to calculate the gain change and a correction value. If the coefficients of the denominators Hv(z) and Huv(z) of the LPC synthesis filters, or the so-called α-parameters, are αi, the characteristics PF(z)of the spectral shaping filter 440 is given by: The fractional part of the equation represents formant stressing characteristics while the portion (1 - kz -1 ) represents high range stressing filter characteristics. In the equation, β, γ and k are constants, such that, for example, β = 0.6, γ = 0.8 and k = 0.3. The gain G of the gain adjustment circuit 443 is given by: in which x(i) and y(i) are an input and an output of the spectral shaping filter 440, respectively. The updating period of the coefficient of the spectral shaping filter 440 is the same as the updating period of the α-parameter which is the LPC synthesis filter coefficient, that is 20 samples or 2.5 msec, whereas the updating period of the gain G of the gain adjustment circuit 443 is 160 samples or 20 msec. By setting the updating period of the gain G of the gain adjustment circuit 443 so as to be longer than that of the coefficient of the spectral shaping filter 440 of the post-filter, it becomes possible to prevent ill effects otherwise caused by gain adjustment fluctuations. Specifically, in a generic post-filter, the updating period of the spectral shaping filter coefficient and the gain updating period are set so as to be equal to each other. If the gain updating period is 20 samples or 2.5 msec, variation occurs within a single pitch period, thus causing click noise. In the present embodiment, the gain switching period is set so as to be longer, that is equal to, for example, 160 samples for one frame, or 20 msec, for preventing gain variations from occurring. Conversely, if the updating period of the spectral shaping filter coefficient is longer, for example, 160 samples or 20 msec, post-filter characteristics cannot follow up with the short-term changes in the speech spectrum, such that the satisfactory psychoacoustic sound quality cannot be achieved. However, more effective post-filtering can be achieved by shortening the filter coefficient updating period to 20 samples or 2.5 msec, For achieving gain connection between neighboring frames, the results of calculations on the filter coefficient and the gain of the previous frame and those of the current frame are multiplied by triangular windows of W(i) = 1/20 (0 ≤ i ≤ 20) and 1 -W(i) (0 ≤ i ≤ 20) for fade-in and fade-out, as shown in Fig.13. Fig.13 shows how the gain G1 of the previous frame is changed to the gain G2 of the current frame. That is, in the overlapping portion, the proportion of the gain and the filter coefficient of the previous frame is decreased gradually while the proportion of the gain and the filter coefficient of the current frame is increased gradually. As for the internal state of the filter at time T, both the filter of the current frame and the filter of the previous frame start from the same state, that is from the last state of the current frame. The above-described signal encoding and signal decoding apparatus may be used as a speech codebook employed in, for example, a portable communication terminal or a portable telephone set shown in Figs.14 and 15. Fig.14 shows a transmitting side of a portable terminal employing a speech encoding unit 160 configured as shown in Figs. 1 and 3. The speech signals collected by a microphone 161 are amplified by an amplifier 162 and converted by an analog/digital (A/D) converter 163 into digital signals which are sent to the speech encoding unit 160 configured as shown in Figs.1 and 3. The digital signals from the A/D converter 163 are supplied to the input terminal 101 of the encoding unit 160. The speech encoding unit 160 performs encoding as explained in connection with Figs.1 and 3. Output signals of output terminals of Figs.1 and 2 are sent as output signals of the speech encoding unit 160 to a transmission channel encoding unit 164 which then performs channel coding on the supplied signals. Output signals of the transmission channel encoding unit 164 are sent to a modulation circuit 165 for modulation and thence supplied to an antenna 168 via a digital/analog (D/A) converter 166 and an RF amplifier 167. Fig.15 shows a reception side of a portable terminal employing a speech decoding unit 260 configured as shown in Figs.2 and 4. The speech signals received by the antenna 261 of Fig.14 are amplified by an RF amplifier 262 and sent via an analog/digital (A/D) converter 263 to a demodulation circuit 264, from which demodulated signals are sent to a transmission channel decoding unit 265. An output signal of the decoding unit 265 is supplied to a speech decoding unit 260 configured as shown in Figs.2 and 4. The speech decoding unit 260 decodes the signals as explained in connection with Figs.2 and 4. An output signal at an output terminal 201 of Figs.2 and 4 is sent as a signal of the speech decoding unit 260 to a digital/analog (D/A) converter 266. An analog speech signals from the D/A converter 266 is sent to a speaker 268. The present invention as defined by the appended claims is not limited to the above-described embodiments. For example, although the structure of the speech analysis side (encoder side) of Figs.1 and 3 or the structure of the speech synthesis side (decoder side) of Figs.2 and 4 are described as hardware, these may also be implemented by a software program using a digital signal processor. On the decoder side, an LPC synthesis filter or a post-filter may be used in common for the voiced speech and the unvoiced speech in place of providing the synthesis filters 236, 237 and the post-filters 238v, 238u as shown in Fig.4. The present invention may also be applied to a variety of usages, such as pitch conversion, speed conversion, computerized speech synthesis or noise suppression, instead of being limited to transmission or recording/reproduction.

Claims: A speech decoding method in which an encoded speech signal is input, decoded and subsequently post-filtered, the method including: performing spectral shaping filtering of the decoded input signal using filter coefficients which are updated periodically; and performing gain adjustment using a gain which is updated periodically for correcting gain changes caused by said spectral shaping filtering, characterised in that the gain is updated with a second period which is longer than the first period with which the filter coefficients are updated. A speech decoding method as claimed in claim 1, wherein said gain is set by comparing the signal level prior to said spectral shaping filtering with that subsequent to said spectral shaping. A speech decoding method as claimed in any one of the preceding claims, wherein said first period is 2.5ms. A speech decoding method a claimed in any one of the preceding claims, wherein said second period is 20ms. A speech decoding apparatus in which an encoded speech signal is input, decoded and subsequently post-filtered, the apparatus including: spectral shaping filtering means (440) for spectrally shaping the decoded input signal with a filter coefficient updated periodically; and gain adjustment means (443, 445) fed with an output of said spectral shaping filtering means and arranged to perform gain adjustment for correcting gain changes caused by said spectral shaping filter means using a gain which is updated periodically, characterised in that the gain adjustment means is arranged to updated the gain with a second period which is longer than the first period with which the filter coefficients are updated. A speech decoding apparatus according to claim 5, wherein said gain is set by comparing the signal level prior to said spectral shaping filtering with that subsequent to said spectral shaping. A speech decoding apparatus as claimed in any one of claims 5 to 6, wherein said first period is 2.5 ms. A speech decoding apparatus as claimed in any one of claims 5 to 7, wherein said second period is 20ms. A portable terminal apparatus comprising: amplifier means (262) for amplifying a reception signal; demodulating means (263) for A/D converting and subsequently demodulating the amplified signal; transmission channel decoding means (265) for channel decoding said demodulated signals; and speech decoding means (260) for decoding an output of said transmission channel decoding means, said speech decoding means including a speech decoding apparatus according to any one of claims 5 to 8; D/A converting means (266) for D/A converting the decoded speech signal for producing an analog speech signal.

Inventors: IIJIMA KAZUYUKI, MATSUMOTO JUN, NISHIGUCHI MASAYUKI, OMORI SHIRO, IIJIMA, KAZUYUKI, MATSUMOTO, JUN, NISHIGUCHI, MASAYUKI, OMORI, SHIRO
Applicants: SONY CORP, SONY CORPORATION
Application year: 1996
Main classification: G10L19
IPC codes: H03M7, G10L19