High-efficiency voltage booster circuit operating at very low supply voltage

Abstract: A voltage booster circuit (30) presenting a pull-up capacitor (48) connected to the supply line (37) via a PMOS switching transistor (46). The other terminal (42) of the pull-up capacitor is supplied with a pull-up voltage (V 1 ) switching between a first value determining charging of the capacitor, and a second value higher than the first and determining pull-up of the capacitor. A negative voltage source (49, 47) presents an output (55) connected to the control terminal of a switch transistor (46), and generates a negative voltage of a value lower than the first pull-up voltage value when charging the capacitor, so as to saturate the switch transistor and charge the capacitor to a voltage close to the supply voltage.

Description: The present invention relates to a high-efficiency voltage booster circuit operating at very low supply voltage. Though applicable in particular to switching regulators featuring N-channel power MOS transistors requiring a high gate potential for them to operate properly, the present invention may also be applied to any device comprising a pass MOS transistor and operating at very low supply voltage. In particular, the present invention may be applied to all MOS and combined BJT/MOS technologies with threshold voltages above or comparable with the supply voltage, e.g. in devices used in pagers, cordless systems, radio-controlled systems and Local Area Networks, supplied by a single low-voltage battery and which, according to present standards, must operate at supply voltages of 800 mV or at most 900 mV. As is known, the physical performance of switching regulators featuring MOS transistors for controlling the current in the inductor should simulate as closely as possible that of an ideal switch. That is, ideally, the MOS transistor should present zero resistance between the drain and source when on. In actual fact, however, this never occurs, and the resistance of the transistor when on equals R ds,on , and must be minimized as far as possible to reduce the voltage drop and power dissipation of the transistor and so achieve a good degree of efficiency of the regulator. As is known, the voltage drop of the transistor is inversely proportional to both its area and overdrive (i.e. the excess gate-source voltage in relation to the turn-on threshold voltage of the transistor), so that, if a limited area of the MOS transistor is required, as in the case of currently used integrated devices, which are necessarily increasingly smaller and mounted in very small packages, the only alternative is to work on the overdrive V ov = V gs - V th . That is, for the switching regulator to operate properly, a high overdrive must be assured. On the other hand, a high overdrive also means a high gate-source voltage V gs which poses problems in the case of circuits with the low supply voltages mentioned previously, in that the minimum threshold voltages achievable by the best technologies currently available are comparable with, if not less than, the low supply voltages required. Nor is it possible, at the low supply voltages considered, to use diodes as unilateral switches. As such, recourse must be made to voltage boosters or pumps capable of generating, from a low supply voltage, a higher voltage for gate driving MOS transistors. Known voltage boosters include what are referred to as Cockroft-Walton second- or higher-order multipliers, which are based on a succession of steps comprising: charging a capacitor to a voltage correlated to the supply voltage; pulling up one terminal of the capacitor; and connecting the other terminal of the capacitor to an output via controlled switches to achieve an output voltage higher than the supply voltage. Figure 1 shows one example of a third-order multiplier 1 comprising three PNP transistors 2-4 and two NPN transistors 5, 6 connected between a supply line at V dd and ground. More specifically, transistor 2 has the emitter terminal connected to the supply, the base terminal connected to the base terminal of transistor 3 and to the emitter terminal of transistor 4, and the collector terminal connected to the collector terminal of transistor 5, to the base terminal of transistor 4 and to the base terminal of transistor 6; transistor 3 has the emitter terminal connected to the supply line, and the collector terminal connected to a node 8; transistor 4 has the collector terminal grounded; transistor 5 has the base terminal defining an input 9 at which it receives a turn-on signal V 1 , and the emitter terminal grounded via a resistor 10; and transistor 6 has the emitter terminal grounded. Circuit 1 also comprises two capacitors 12, 13, one terminal 15, 16 of which defines an input for respective push-pull digital control signals S1, S2, and the other terminal A, B of which is connected to a respective terminal of a chain of diodes 17, 18, 19. More specifically, diode 17 is connected at the anode to node 8, and at the cathode to terminal A and to the anode of diode 18; the cathode of diode 18 is connected to terminal B and to the anode of diode 19; and the cathode of diode 19 forms an output terminal 20 presenting voltage O. Components 2-6 and 10 form an input stage for transferring the supply voltage to node 8 when V 1 is high. In this condition, it is assumed S1 is low and S2 high at instant t 0 , so that potential V A at terminal A equals: V A (t 0 ) = V dd - V cesat,3 - V d,17 where V cesat,3 is the collector-emitter voltage drop of transistor 3 when saturated; and V d,17 is the voltage drop of diode 17. Potential V B at terminal B, on the other hand, equals: V B (t 0 ) = V dd - V cesat,3 - V d,17 - V d,18 where V d,18 is the voltage drop of diode 18. For example, if voltage V dd equals 5 V, if V cesat,3 equals 0.3 V, and if the voltage drop of the diodes equals 0.7 V, then V A (t 0 ) = 4 V and V B (t 0 ) = 3.3 V. At instant t 1 , S1 switches to high (V dd ) and S2 to low, and, since capacitor 12 is already charged, terminal A is brought to potential: V A (t 1 ) = V A (t 0 ) + V dd diode 17 gets off, and terminals B and 20 are brought to potential: V B (t 1 ) = V A (t 1 ) - V d,18 V 20 (t 1 ) = V B (t 1 ) - V d,19 where V d,19 is the voltage drop of diode 19. In the example shown, V A (t 1 ) = 9 V, V B (t 1 ) = 8.3 V and V 20 (t 1 ) = 7.6 V. At instant t 2 , S1 switches back to low and S2 to high (V dd ), diode 18 gets off, and the potentials at terminals A, B and 20 equal: V A (t 2 ) = V A (t 0 ) = 4 V V B (t 2 ) = V B (t 1 ) + V dd = 13.3 V V 20 (t 2 ) = V B (t 2 ) - V d,19 = 12.6 V As can be seen, the loss in efficiency of the Figure 1 circuit is considerable, mainly on account of the voltage drop of diodes 17-19, and prevents it from being used in the presence of very low supply voltages (800 mV mentioned previously) at which it is not even possible to charge capacitors 12, 13 on account of the voltage drop of diodes 17, 18. A voltage booster according to the preamble of claim 1 is disclosed e.g. in US-A-5,162,688 describing a voltage regulator having a charge pump. The charge pump of this document suffers the same disadvantages described above for the circuit of figure 1. It is an object of the present invention to provide a voltage booster circuit capable of operating even with a low supply voltage, and in particular when the supply voltage is comparable with or even below the threshold voltage of MOS transistors. According to the present invention, there is provided a high-efficiency voltage booster circuit with a very low supply voltage, as claimed in Claim 1. In practice, according to the present invention, the pull-up capacitor is charged by a MOS transistor which, when saturated, presents a very low drain-source voltage drop, and so permits the capacitor to be charged almost to the supply voltage; and, to ensure a gate-source voltage sufficient to saturate the MOS transistor even with a very low supply voltage, a P-channel transistor is used, and a negative gate drive voltage is generated when charging the capacitor. A preferred, non-limiting embodiment of the present invention will be described by way of example with reference to the accompanying drawings, in which: Figure 1 shows an electric diagram of a known voltage booster; Figure 2 shows a circuit diagram of one embodiment of the voltage booster according to the present invention; Figures 3 and 4 show plots of voltages measured at given points in the Figure 3 circuit; Figure 5 shows a cross section of a wafer of semiconductor material implementing a P-channel MOS transistor; Figure 6 shows a variation of the Figure 2 diagram; Figure 7 shows an electric diagram of a third-order voltage booster in accordance with the present invention; Figure 8 shows plots of voltages measured in the Figure 7 circuit. Number 30 in Figure 2 indicates a voltage booster circuit comprising an input stage 31, a charge pump stage 32, and an output stage 33. Input stage 31 is a bipolar inverting type comprising a first PNP bipolar transistor 35 and a second NPN bipolar transistor 36, both connected between a supply line 37 at V dd and a ground line 38. More specifically, transistor 35 has the emitter terminal connected to supply line 37, the base terminal connected to an input terminal 39 via a resistor 40, and the collector terminal connected to a node 42; transistor 36 has the collector terminal connected to node 42, the base terminal connected to input terminal 39 via a resistor 41, and the emitter terminal connected to ground 38; and input terminal 39 receives a square-wave signal OSC, not necessarily rail-to-rail, generated by an oscillator (not shown). Charge pump stage 32 comprises a third and fourth bipolar transistor 44, 45, a first and second MOS transistor 46, 47, and a first and second capacitor 48, 49. Transistor 44, which is a PNP type, has the emitter terminal connected to supply line 37, the base terminal connected to input terminal 39 via a resistor 51, and the collector terminal connected to a node 55; transistor 47, which is an N-channel type, has the drain terminal connected to node 55, the gate terminal connected to input terminal 39, the source terminal connected to a node 56, and the body region connected to the source terminal; transistor 45, which is an NPN type, has the collector terminal connected to node 56, the base terminal connected to node 42 via a resistor 52, and the emitter terminal grounded; and a parasitic PNP transistor 53 is also shown by the dotted line, with its collector in common with the base region of transistor 45, the base in common with the collector region of transistor 45, and the emitter grounded. Transistor 46, which is a P-channel type, has the source terminal connected to supply line 37, the gate terminal connected to node 55, the drain terminal connected to a node 57, and the body region connected to the source terminal; capacitor 48 has a first terminal connected to node 57, and a second terminal connected to node 42; and capacitor 49 has a first terminal connected to node 42, and a second terminal connected to node 56. Output stage 33 comprises a P-channel MOS transistor 60 and a capacitor 61. Transistor 60 has the source terminal connected to node 57, the gate terminal connected to input terminal 39, the drain terminal connected to an output terminal 62, and the body connected to the drain terminal; capacitor 61 is connected between output terminal 62 and ground 38; and output terminal 62 may be connected to the gate terminal of an NMOS transistor (not shown) forming part of a switching regulator as stated previously. Operation of the Figure 2 circuit will be described with reference also to Figures 3 and 4. Initially, when the circuit is turned on, voltage V dd is low but rising, and input signal OSC is low, so that transistors 36 and 47 are off, and transistors 35 and 44 are turned on. During the transient state in which supply voltage V dd is increasing (after about 25 µ s in the simulation shown), a voltage peak occurs at node 56 (voltage V 3 in Figure 3) due to the low OSC signal. In fact, transistor 35 is conducting and pulls up node 42, and capacitor 49, which is discharged, connects node 56 to node 42. As supply voltage V dd increases, transistor 45 is also turned on and for the same area, conducts three times the current of transistor 44. As a result, transistor 45 draws node 56 towards ground and down to the saturation voltage of transistor 45, so that capacitor 49 is charged to voltage V 1 - V 3 according to the equation: V 1 - V 3 = V dd - V cesat,35 - V cesat,45 where V cesat,35 and V cesat,45 are the collector-emitter saturation voltages of transistors 35, 45 respectively. At the end of the output transient state of supply voltage V dd (after about 30 µ s), input signal OSC switches to high, thus turning off transistors 35, 44 and turning on transistor 36, which saturates and turns off transistor 45. As a result, node 42, at potential V 1 , falls to the saturation voltage V cesat,36 of transistor 36, and, having been charged during the turn-on transient state, capacitor 49 pushes node 56 to a negative potential V 3 according to the equation: V 3 = (V dd - V cesat,35 - V cesat,45 - V cesat,36 ) as shown in Figure 3. In the Figure 3 simulation using a V dd of 800 mV, V 3 presents a negative swing to about -600 mV. Together with the high state of signal OSC, this negative voltage produces, between the gate and source terminals of MOS transistor 47, a voltage drop V gs,47 of: V gs,47 = OSC + V 3 As a result, assuming OSC in the high state presents a value of 740 mV (peak-peak voltage of 680 mV), then V gs,47 = 1.34 V, which is more than sufficient to turn on and saturate transistor 47. The extremely small drain-source voltage drop V ds,47 and the off state of transistor 44 connect the gate terminal of P-channel transistor 46 to -600 mV voltage V 3 , so that, disregarding voltage V ds,47 , transistor 46 presents a gate-source voltage drop V gs,46 according to the equation: V gs,46 = V dd - V 3 ≈ 1.4 V The turning on of transistor 46 brings node 57 to voltage V 2 ≈ V dd = 800 mV, and causes capacitor 48 to charge to voltage V 2 - V 1 according to the equation: V 2 - V 1 = V dd - V ds,46 - V cesat,36 ≈ 700 mV As such, transistor 60 of output stage 33 is off, due to the high potential (OSC) at its gate terminal, but nevertheless conducts a small amount of current, as explained below with reference to Figure 5 showing a simplified cross section of a P-channel MOS transistor. As shown in Figure 5, transistor 60 (or 46, which is formed the same way) comprises an N-type body region 70 housing a P-type source region 71 and a P-type drain region 72 separated by a portion of body region 70 forming channel 73; a gate region 74 is located over channel region 73 and is insulated electrically from it by an oxide layer 75; and source and gate regions 71, 72 form PN junctions with body region 70 (diodes 77, 78 shown by the dotted lines). As shown schematically, and unlike standard MOS transistors, body region 70 is shorted to drain region 72. Due to the peculiar connection of body region 70 of transistor 60 to drain region 72, and the fact that capacitor 61 is still at a low voltage (as in the first cycle of the OSC signal), parasitic diode 77 between source region 71 (connected to node 57 at V dd ) and body region 70 (connected to output terminal 62 still at a low voltage) is biased directly and conducts a small amount of current, so that capacitor 61 begins charging, as shown in the Figure 7 simulation. On switching back to low, signal OSC turns transistors 47, 36 off and transistors 35, 44 on, so that transistor 46 is also turned off and transistor 45 on to increase potential V 3 at node 56 once more to V cesat,45 . The simultaneous rise in potential V 1 at node 42, by virtue of being connected to supply line 37 by transistor 35, on the one hand charges capacitor 49 for the next cycle, and on the other increases potential V2 at node 57 to: V 2 = (V dd - V ds,46 - V cesat,36 ) + (V dd - V cesat,35 ) where the first term in brackets is due to the previous charge of capacitor 48, and the second term in brackets to potential V 1 at node 42. Since, at this point, the potential at the gate terminal of transistor 60 is low (like signal OSC) and its source terminal (connected to node 57) is at the above potential V 2 , the gate-source voltage drop is high enough to make transistor 60 highly conductive and so permit charge transfer from capacitor 48 to capacitor 61, as shown by the V 2 and V 0 curves in the Figure 3 simulation. Alongside successive switching of signal OSC, capacitor 61 is charged up to about 1.4 V, as shown in the Figure 4 simulation with a smaller time scale than in Figure 3. As can be seen, with the numerical values indicated, output voltage V 0 therefore reaches a value of 1.4 V to be compared to the maximum theoretical value of 2V dd = 1.6 V, so that the circuit presents an efficiency of 88.125% and is capable of functioning with a supply voltage V dd of as little as 700 mV or even 600 mV. In the Figure 2 circuit, resistors 40,41 and 51,52 provide for introducing a certain delay between the switching of signal OSC and transistors 35,36 and 44,45 respectively, to ensure synchronous operation of the circuit as a whole, which is further assisted by transistor 47 ensuring immediate latching of node 55 to node 56, and rapid negative voltage transition of node 56 following switching of signal OSC. The parasitic transistor 53 shown by the dotted line in Figure 2 is turned on when node 56 falls below ground in the case of a P-type substrate. In fact, together with the P-type substrate, the base-collector junction of transistor 45 forms a parasitic vertical PNP transistor having the base of transistor 45 as the collector, the collector of transistor 45 as the base, and the grounded substrate as the emitter. When node 56 (as also the collector of 45 and the base of 53) falls below ground, the base terminal of transistor 45 (collector of 53) is connected to node 42 slightly above ground, and directly biases the base-collector junction of transistor 53, which is turned on and limits the fall of node 56 at potential V 3 to its base-emitter voltage drop V be , as shown in the Figure 3 simulation. Even with this limitation, the circuit nevertheless operates correctly as described above. The peculiar connection of the body region of transistors 46 and 61 in the Figure 2 circuit will now be explained with reference to Figure 5. If the body region 70 of transistor 46 were shorted to source region 77 in the usual way, problems could arise when turning off transistor 46, when signal OSC is high and node 57 is at a much higher voltage than V dd . In this case, in fact, diode 78 at the drain-body junction would be biased directly and turned on, thus maintaining the voltage drop at the junction equal to V be . As a result, the diode would limit the increase in potential V 2 at node 57 to V dd + V be , i.e. to the potential of the source and body regions plus the voltage drop of the diode, as well as discharging capacitor 48 towards body region 70. Conversely, with the connection shown, this does not occur. In fact, parasitic diode 78 is shorted, and parasitic diode 77 between the body and source regions is biased inversely when transistor 46 is off (body region 70 at a much higher voltage V 2 than V dd , and source region 71 at V dd ), and is biased at below threshold voltage when transistor 46 is on (body region 70 at a lower voltage than source region 71, to the amount of V ds,46 equal to a few mV). Similarly, the peculiar connection of the body region of transistor 60 has the effect of charging capacitor 61 as soon as the circuit is turned on, even when transistor 60 is off, as described above. This, however, is in no way disadvantageous, and the connection provides for ensuring correct operation after capacitor 61 is charged. In fact, as shown in Figures 2 and 5, a conventional connection of body region 70 to source region 71 would turn on diode 78 between the drain and body when transistor 60 is off. More specifically, in this condition (signal OSC high, capacitor 48 charging), the potential of the drain region (connected to capacitor 61 charged to almost 2V dd ) would be much higher than the potential of source region 71 and body region 70 (roughly equal to V dd ) and at any rate higher than the threshold voltage of diode 78, so that diode 78, when on, would discharge capacitor 61 towards node 57. Conversely, with the connection shown, diode 78 is shorted, and diode 77 is biased inversely when transistor 60 is off. Should circuit 30 in Figure 2 be operated at supply voltages higher than the low levels considered (V dd of 1.2-1-3 V and over), when charging capacitor 48 (when signal OSC is high and transistor 60 should be off), the drain-gate voltage drop of transistor 60 is greater than its turn-on threshold, so that it can never be turned off, thus discharging output capacitor 61 towards stage 32. To solve this problem, instead of being controlled directly by square-wave signal OSC, transistor 60, in circuit 80 in Figure 6, is controlled by a P-channel MOS transistor 82 and a bipolar NPN transistor 83. More specifically, the gate terminal of transistor 60 is connected to a node 85 also connected to the drain terminal of transistor 82 and the collector terminal of transistor 83; transistor 82 has the source terminal connected to output terminal 62, the gate terminal connected to node 55, and the substrate connected to the source terminal; and transistor 83 has the emitter terminal connected to line 38 (ground), and the base terminal connected to node 42 via a resistor 86. The rest of circuit 80 is identical to circuit 30 in Figure 2, and therefore requires no further description. When signal OSC switches to low, therefore, node 42 switches to high and (with a slight delay due to resistor 86) turns on transistor 83, which grounds node 85, thus turning on transistor 60 which operates in exactly the same way as in the Figure 2 circuit, to permit charge transfer from capacitors 48, 49 to capacitor 61. At this stage, transistor 82 is off, due to the high state of node 55. Conversely, when signal OSC switches to high and node 55 is below ground due to capacitor 49 and transistor 47, transistor 82 is turned on and shorts the drain and gate terminals of transistor 60 to ensure that transistor 60 is turned off. At this stage, transistor 83 is off. Figure 7 shows a variation of the Figure 6 circuit operating as a voltage triplicator, i.e. as a Cockroft-Walton third-order multiplier. Instead of being connected directly to output terminal 62, the drain terminal of transistor 60 in circuit 90 in Figure 7 is connected via a P-channel MOS transistor 92 controlled by an inverter 93. More specifically, transistor 92 has the source terminal (defining node 94) connected to the drain terminal of transistor 60, the gate terminal connected to output node 95 of inverter 93, the drain terminal connected to output terminal 62 of the circuit, and the substrate connected to the drain terminal; inverter 93 comprises a pair of complementary transistors 96, 97; transistor 96, which is a PMOS type, has the source terminal connected to output terminal 62 (and to its own substrate), the drain terminal connected to output node 95, and the gate terminal connected to node 85; and transistor 97, which is an NMOS type, has the drain terminal connected to node 95, the gate terminal connected to node 85, and the source terminal (and substrate) connected to ground line 38. Circuit 90 also comprises a capacitor 98 connected between node 94 and an input terminal 99 supplied with signal OSC at terminal 39. The rest of circuit 90 is identical to circuit 80. In circuit 90 in Figure 7, when signal OSC is low, node 57 is at roughly 2V dd (as described with reference to Figure 2); node 42 is high; transistor 83 is turned on; node 85 is practically grounded; transistor 60 is turned on and also brings node 94 to practically 2V dd ; terminal 99 is also low, so that capacitor 98 is charged to said potential; and, node 85 being grounded, transistor 96 is turned on, thus shorting the drain and gate terminals of transistor 92 which is therefore off. When signal OSC switches to high, transistor 47 is turned on, so that node 55 is brought below ground, as described previously; transistor 82 is turned on and turns transistor 60 off; node 94 is brought to roughly 3V dd due to the high state of input terminal 99; transistor 82 brings node 85 to roughly 3V dd , thus turning off transistor 96 and turning on transistor 97 which grounds node 95; and transistor 92 is turned on to latch node 94 to output terminal 62 and so permit charge transfer from capacitor 98 to capacitor 61. Operation of the Figure 7 circuit is also clearly deducible from the Figure 8 simulation, wherein V 1 indicates the voltage at node 42, V 2 the voltage at node 57, V 4 the voltage at node 95, V 5 the voltage at node 85, V 6 the voltage at node 94, and V 0 the output voltage at terminal 62. During the first few operating cycles, the efficiency of circuit 90 is poor, due to the uncertainty of transistor 96, which is not perfectly turned on when charging capacitor 98 (signal OSC low) or perfectly turned off at the opposite stage. In fact, initially, when output voltage V 0 at output terminal 62 is still low (V 0 < V th , where V th is the threshold voltage of transistor 96) and signal OSC and node 85 are low, the source-gate voltage drop of transistor 96 is also low, thus preventing it from being turned on properly and in turn from ensuring transistor 92 is turned off altogether. Similarly, when signal OSC is high, transistor 96 may not be turned off altogether, thus preventing transistor 92 from being turned on fully. This does not prevent the circuit from being started correctly, however. In fact, when signal OSC is low, on account of the peculiar connection of the substrate and drain of transistor 92, as described with reference also to Figure 5, diode 77 between the source region and body/drain region of transistor 92 is biased directly and turned on, and therefore connects the potential of node 94 to output terminal 62 to permit a rapid initial charge of capacitor 61 until output voltage V 0 reaches the threshold value of transistor 96. Conversely, when the output voltage is high (<2V dd ), and it is essential, when signal OSC is low, that transistor 92 be turned off to prevent return of the charge, transistor 96 is definitely turned on. The advantages of the circuit described are as follows. Firstly, as stated, by virtue of featuring no diodes or bipolar transistors as switching elements, the circuit according to the invention presents a high degree of efficiency, even in the case of low-capacitance pull-up capacitors. Secondly, by virtue of clipper-inverter stage 31, which also operates as a level shifter and delay controller, the circuit may be driven by a square-wave, non-rail-to-rail clock voltage. And thirdly, by virtue of the circuit arrangement shown, an exponential increase in supply voltage may be achieved using a number of cascade-connected circuits, as shown in Figure 7, to form a third-order multiplier and so enable troublefree control of the drive voltages at the gate terminals of power MOS transistors. Clearly, changes may be made to the circuit as described and illustrated herein without, however departing from the scope of the present invention as claimed. In particular, in parallel with bipolar transistors 35, 36, 44, 45 and possibly also 83, provision may be made for respective MOS transistors capable of superimposing their perfect-switch characteristic (low R ds,on ) on that of the bipolar transistors to enhance the efficiency of the circuit, providing the supply voltage V dd and the amplitude of signal OSC exceed the MOS threshold voltages in the technology employed. In which case, the bipolar transistors in parallel with the MOS transistors will only conduct minimally, the current being almost completely drained by the MOSFET transistors.

Claims: A voltage booster circuit (30) comprising a first reference potential line (37) set at a first reference potential (Vdd); at least a first pull-up capacitor (48) with a first (57) and second (42) terminal; and a PMOS switching transistor (46) with a control terminal (46), said PMOS switching transistor being connected between said first reference potential line (37) and said first terminal (57) of said first capacitor (48); said second terminal (42) of said first capacitor receiving a pull-up signal (V1) switching between a first value and a second value higher than the first value and near to said first reference potential (Vdd); characterized by negative voltage source means (49, 47) having an output (55) connected to said control terminal of said switching transistor (46), for generating a control signal of a third value lower than said first value of said pull-up signal; said negative voltage source means being turned on when said pull-up signal presents said first value. A circuit as claimed in Claim 1, characterized in that said negative voltage source means (49, 47) comprise a second capacitor (49) with a first and second terminal; and a MOS latch transistor (47) with a first, second and control terminal; said first terminal of said second capacitor (49) being connected to said second terminal (42) of said first capacitor (48); said second terminal of said second capacitor being connected to said first terminal (56) of said MOS latch transistor (47); said second terminal of said MOS latch transistor being connected to said control terminal (55) of said switching transistor (46); and said control terminal of said latch transistor receiving a square-wave oscillating signal (OSC). A circuit as claimed in Claim 2, characterized in that it comprises an inverting stage (31) interposed between said first reference potential line (37) and a second reference potential line (38); said inverting stage presenting an input (39) receiving said square-wave oscillating signal (OSC), and an output (42) connected to said second terminal of said first capacitor (48) and to the control terminal of a bipolar switch (45); and said bipolar switch being interposed between said second terminal (56) of said second capacitor (49) and said second reference potential line (38). A circuit as claimed in Claim 3, characterized in that said inverting stage (31) comprises a first (35) and second (36) bipolar transistor of opposite types, and each presenting a first, second and control terminal; said first terminal of said first bipolar transistor (35) being connected to said first reference potential line (37); said second terminal of said second bipolar transistor (36) being connected to said second reference potential line (38); said control terminals of said first and second bipolar transistors being connected to said input (39) of said inverting stage (31); and said second terminal of said first bipolar transistor and said first terminal of said second bipolar transistor being connected to each other and defining said output (42) of said inverting stage. A circuit as claimed in Claim 4, characterized in that it comprises a first (40) and second (41) resistor interposed between said input (39) of said inverting stage (31) and respective control terminals of said first and second bipolar transistors (35, 36). A circuit as claimed in Claim 4 or 5, characterized in that said bipolar switch comprises a third bipolar transistor (45); and in that it comprises a fourth bipolar transistor (44) with a first terminal connected to said first reference potential line (37), a second terminal connected to said control terminal of said MOS switching transistor (46), and the control terminal connected to said input (39) of said inverting stage (31). A circuit as claimed in Claim 6, characterized in that it comprises a third (52) and fourth (51) resistor; said third resistor (52) being interposed between said output (42) of said inverting stage (31) and said control terminal of said third bipolar transistor (45); and said fourth resistor (51) being interposed between said input (39) of said inverting stage (31) and said control terminal of said fourth bipolar transistor (44). A circuit as claimed in Claim 6 or 7, characterized in that said first and fourth bipolar transistors (35, 44) are PNP types; said second and third bipolar transistors (36, 45) are NPN types; and said MOS latching transistor (47) is an N-channel type. A circuit as claimed in any one of the foregoing Claims from 3 to 8, characterized in that it comprises an output stage (33) including a first MOS output transistor (60) and an output capacitor (61) in series with each other between said first terminal (57) of said first capacitor (48) and said second reference potential line (38). A circuit as claimed in Claim 9, characterized in that said first MOS output transistor (60) is a P-channel type with its control terminal connected to said input (39) of said inverting stage (31). A circuit as claimed in Claim 9, characterized in that said first MOS output transistor (60) is a P-channel type with its control terminal connected to a first intermediate node (85) formed by the series connection of a first and second controlled switch (82, 83); said first controlled switch (82) being interposed between said first intermediate node (85) and the drain terminal of said first MOS output transistor (60), and having its own control terminal connected to said second terminal of said MOS latching transistor; and said second controlled switch (83) being interposed between said first intermediate node (85) and said second reference potential line (38), and having its own control terminal connected to said output of said inverting stage (31). A circuit as claimed in any one of the foregoing Claims from 9 to 11, characterized in that said MOS switch transistor (46) and said first MOS output transistor (60) comprise a body region (70) and a drain region (72) shorted to each other. A circuit as claimed in any one of the foregoing Claims from 9 to 12, characterized in that it comprises a voltage triplicating stage comprising a second MOS output transistor (92) with a first and second terminal respectively connected to said first MOS output transistor (60) and said output capacitor (61); a third capacitor (98) with a first terminal receiving said square-wave oscillating signal, and a second terminal connected to said first terminal of said second MOS output transistor (92); and an inverting circuit (93) comprising a third and fourth controlled switch (96, 97) complementary to each other and defining a second intermediate node (95); said third and fourth controlled switches having the control terminals connected to each other and receiving a switching signal synchronous with said square-wave oscillating signal; said second intermediate node (95) being connected to the control terminal of said second MOS output transistor (92); said third controlled switch (96) being connected between said second terminal and said control terminal of said second MOS output transistor; and said fourth controlled switch (97) being connected between said second intermediate node (95) and said second reference potential line (38).

Inventors: CRISCIONE MARCELLO, SCILLA GIUSEPPE, CRISCIONE, MARCELLO, SCILLA, GIUSEPPE
Applicants: CONS RIC MICROELETTRONICA, CONSORZIO PER LA RICERCA SULLA
Application year: 1996
Main classification: H02M3
IPC codes: H02M3